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in NBS TECHNICAL NOTE 652

U.S. DEPARTMENT OF COMMERCE/National Bureau of Standards

Development and Construction

of an Electromagnetic

Near-Field Synthesizer
NATIONAL BUREAU OF STANDARDS

The National Bureau of Standards' was established by an act of Congress March 3. 1901. The Bureau's overall goal is to strengthen and advance the Nation's science and technology

and facilitate their effective application for public benefit. To this end, the Bureau conducts

research and provides: (I) a basis for the Nation's physical measurement system, (2) scientific and technological services for industry and government, ( 3 ) a technical basis for equity in trade, and (4) technical services to promote public safety. The Bureau consists of the Institute for Basic Standards, the Institute for Materials Research, the Institute for Applied Technology,

the Institute for Computer Sciences and Technology, and the Office for Information Programs.

THE INSTITUTE FOR BASIC STANDARDS provides the central basis within the United

States of a complete and consistent system of physical measurement; coordinates that system

with measurement systems of other nations: and furnishes essential services leading to accurate and uniform physical measurements throughout the Nation's scientific community, industry, and commerce. The Institute consists of a Center for Radiation Research, an Office of Measurement Services and the following divisions: Applied Mathematics — Electricity — Mechanics — Heat — Optical Physics — Nuclear

Sciences 2 — Applied Radiation = — Quantum Electronics * — Electromagnetics 8 — Time

and Frequency " — Laboratory Astrophysics ' — Cryogenics ". THE INSTITUTE FOR MATERIALS RESEARCH conducts materials research leading to improved methods of measurement, standards, and data on the properties of well-characterized

materials needed by industry, commerce, educational institutions, and Government; provides

advisory and research services to other Government agencies; and develops, produces, and

distributes standard reference materials. The Institute consists of the Office of Standard Reference Materials and the following divisions: Analytical Chemistry — Polymers — Metallurgy — Inorganic Materials — Reactor

Radiation — Physical Chemistry.

THE INSTITUTE FOR APPLIED TECHNOLOGY provides technical services to promote

the use of available technology and to facilitate technological innovation in industry and Government; cooperates with public and private organizations leading to the development of

technological standards (including mandatory safety standards), codes and methods of test; and provides technical advice and services to Government agencies upon request. The Institute consists of a Center for Building Technology and the following divisions and offices: Engineering and Product Standards — Weights and Measures — Invention and Innovation — Product Evaluation Technology — Electronic Technology — Technical Analysis — Measurement Engineering — Structures, Materials, and Life Safety * — Building

Environment ' — Technical Evaluation and Application * — Fire Technology.

THE INSTITUTE FOR COMPUTER SCIENCES AND TECHNOLOGY conducts research and provides technical services designed to aid Government agencies in improving cost effec- tiveness in the conduct of their programs through the selection, acquisition, and effective utilization of automatic data processing equipment; and serves as the principal focus within

the executive branch for the development of Federal standards for automatic data processing

equipment, techniques, and computer languages. The Institute consists of the following

divisions: Computer Services — Systems and Software — Computer Systems Engineering — Information Technology.

THE OFFICE FOR INFORMATION PROGRAMS promotes optimum dissemination and

accessibility of scientific information generated within NBS and other agencies of the Federal Government; promotes the development of the National Standard Reference Data System and

a system of information analysis centers dealing with the broader aspects of the National Measurement System; provides appropriate services to ensure that the NBS staff has optimum

accessibility to the scientific information of the world. The Office consists of the following

organizational units: Office of Standard Reference Data — Office of Information Activities — Office of Technical

Publications — Library — Office of International Relations.

1 Headquarters and Laboratories at Gaithersburg, Maryland, unless otherwise noted; mailing address Washington, D.C. 20234.

2 Part of the Center for Radiation Research.

3 Located at Boulder. Colorado 80302.

* Part of the Center for Building Technology.
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CO,'! 1- 1974

753

Development and Construction of an

Electromagnetic Near-Field Synthesizer

Frank M. Greene

Electromagnetics Division

Institute for Basic Standards

U'S. National Bureau of Standards

Boulder, Colorado 80302

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U.S. DEPARTMENT OF COMMERCE, Frederick B. Dent, Secretary

NATIONAL BUREAU OF STANDARDS. Richard W Roberts. Director Issued May 1974
National Bureau of Standards Technical Note 652

Nat. Bur. Stand. (U.S.), Tech. Note 652, 44 paget (May 1974)

CODEN: NBTNAE

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402

(Order by SD Catalogue No. 03.46:652) $0.50
CONTENTS

Page

1. INTRODUCTION 2

2. THE RF MAGNETIC -FIELD GENERATOR 3

2.1 The Physical Characteristics of the Loop Inductor-- 3

2.2 The Electrical Characteristics of the Loop Inductor 4

2.3 The Loop- Inductor Matching Networks 9

2.4 The Vacuum Variable Tuning Capacitors 11

2.5 The Balun Transformer 11

2.6 The Water Cooling of the Inductor 12

3. THE RF ELECTRIC-FIELD GENERATOR 15

3.1 The Physical Characteristics of the Parallel -Plate

Strip Line 15

3.2 The Parallel-Plate Matching Network 16

3.3 The Electrical Characteristics of the ParallelPlate Strip Line 19

4. THE COMPLETE SYNTHESIZER UNIT 19

4.1 A General Description 19

4.2 The Effect of the Shielded Room 20

4.3 The RF Excitation of the Field Generators 21

4.4 The Phase-Shift Networks 21

4.5 The RF Exposure Capability 23

5. SUMMARY AND CONCLUSIONS 24

6. ACKNOWLEDGMENT 2 6

APPENDIX A

7. THE ELECTRIC FIELD ASSOCIATED WITH A LOOP INDUCTOR 27

7.1 Introduction 27

7.2 The Single-Source Loop Inductor 28

7.3 A Double -Source Loop Inductor 29

7.4 A Quadruple -Source Loop Inductor 30

7.5 A Loop Inductor with a Continuously-Distributed

Source 31

APPENDIX B

8. METHODS USED TO SIMULATE THE MULTIPLE SOURCES 3 3

8.1 Introduction 33

8.2 The Shielded Half-Loop with a Gap at One End 33

8.3 The Shielded Half-Loop with a Gap at the Center 35

8.4 The Shielded Quarter-Loop with a Gap at the Center- 35

8.5 Calculation of the Loop Characteristics 37

9. REFERENCES 3 9

tit
LIST OF FIGURES

Page

Fig. 1 Diagram of the Magnetic-Field Generator -- 6

Fig. 2 View of the Four-Gap Loop Inductor 7

Fig. 3 View of the Inductor and Matching Networks 8

Fig. 4 Circuit Diagram of the Matching Network 9

Fig. 5 View of the Matching Network 10

Fig. 6 View of the Toroidal Balun Transformer 13

Fig. 7 A Basic Balun Circuit 14

Fig. 8 Winding the Toroidal Balun Transformer 14

Fig. 9 Diagram of the Near-Field Synthesizer 16

Fig. 10 View of the Near-Field Synthesizer 17

Fig. 11 Circuit Diagram of the Matching Network 18

Fig. 12 Diagram of the Complete Synthesizer 22

Fig. 13 Diagram of the Phase-Shift Network- 23

APPENDIX A

Fig. 14 Single -Source E-Field Distribution 29

Fig. 15 Double -Source E-Field Distribution 30

Fig. 16 Quadruple -Source E-Field Distribution 30

Fig. 17 Continuous -Source E-Field Distribution 32

APPENDIX B

Fig. 18 Half-Loop with a Gap at One End 34

Fig. 19 Half-Loop with a Gap at the Center 36

Fig. 20 Quarter-Loop with a Gap at the Center 36

IV
DEVELOPMENT AND CONSTRUCTION OF AN ELECTROMAGNETIC

NEAR-FIELD SYNTHESIZER

Frank M. Greene

ABSTRACT

This publication describes work done by the National

Bureau of Standards for the USAF School of Aerospace Medicine

at Brooks AF Base involving the development, design, construction and testing of a prototype EM near-field synthesizer. The

purpose of the contract was to provide a means of independently

generating high-level electric and magnetic near fields in the

frequency range 10 to 30 MHz. These fields are to be used in

various ratios by the USAFSAM in their EM radiation exposure

program for determining the biological effects of hazard-level

non-ionizing EM fields on human beings.

The synthesizer consists of a balanced, parallel -plate

strip line to generate the "desired" electric field, and a single-turn quadruple -feed inductor placed parallel to and midway between the plates to generate the "desired" magnetic

field. Methods used to reduce the "unwanted" E- and H-field

components associated with the above, as well as the methods

used to reduce the coupling between the two field systems are

discussed. The result is a synthesizer in which the electricand magnetic- field components can be adjusted essentially independently over wide ranges of magnitude, relative time -phase,

and spatial orientation to simulate various near-field configur

at ions

.

Previous research has been largely limited to the use of

plane-wave fields for evaluating RF biological hazards. This

new device will allow researchers to investigate any near-field

effects that may occur at high field levels.

Keywords: Electromagnetic-field hazards; electromagneticfield synthesizer; electroma.gne tic radiationexposure testing (non -ionizing) ; near fields;

RF biological hazards.
1. INTRODUCTION

This publication describes work done by the National

Bureau of Standards for the USAF School of Aerospace Medicine

at Brooks Air Force Base involving the development, design,

construction, and testing of a prototype EM near-field synthesizer. The purpose of the contract was to provide a means of

independently generating high-level electric and magnetic

near-fields in the frequency range 10 to 30 MHz. These fields

are to be used in various ratios by the USAFSAM in their EM

radiation exposure program for determining the biological

effects of hazard-level non-ionizing EM fields on human beings.

The synthesizer consists of: (a) a balanced, parallelplate strip line to generate the "desired" electric field;

and (b) a single -turn quadruple -feed inductor placed parallel

to and midway between the plates to generate the "desired"

magnetic field. It is generally known from Maxwell's equations

that in addition to the "desired" E and H fields in (a) and (b)

respectively, the electric field in (a) is accompanied by an

"unwanted" magnetic field, and the magnetic field in (b) is

accompanied by an "unwanted" electric field.

It turns out that the "unwanted" magnetic field produced

by the parallel -plate strip line is negligibly small for the

plate size, frequency, and the magnitude of E being used. It

will be shown how the magnitude of the "unwanted" E-field produced by the single -turn inductor was reduced by an order of

magnitude in the central portion of the inductor by the expediency of using a mult iple -feed rather than the conventional

single feed. This means that under the conditions of use the

single-turn inductor can generate a fairly "pure," high-level

H-field, and the parallel -plate line a fairly "pure," highlevel E-field over the above frequency range.
The electric and magnetic coupling between the two field

systems was minimized by operating each system in a balanced

mode and with the required mutual symmetry. There is no

observable cross-coupling even when the loop is rotated out

of its initial plane of symmetry by any desired angle. As a result, the electric- and magnetic-field components can be

adjusted essentially independently over wide ranges of magnitude, relative electrical time -phase, and relative spatial

orientation in order to simulate various near-field configurations. The maximum RF driving power required is one

kilowatt for each field system.

Previous research in RF biological hazards has been

largely limited to the use of plane -wave fields for making

clinical studies. This new device will allow researchers

to investigate any near-field effects that may occur at highfield levels. The practical mks International System of Units

(SI) is used throughout this paper.

2. THE RF MAGNETIC -FIELD GENERATOR

2.1 The Physical Characteristics of the Loop Inductor . The

RF magnetic-field generator consists basically of a singleturn inductor having an average diameter of 21 5/8" (54.9 cm)

formed from copper tubing having an outside diameter of 1 5/8"

(41.3 mm) and a wall thickness of 1/16" (1.59 mm). The inductor has a low-frequency inductance of 0.921 microhenry as

determined from Equation (2) in Appendix A. The diameter

selected was the result of a compromise to make the inductor

as small as possible (to maintain a reasonably high selfresonant frequency) while still providing sufficient space to

accomodate the primate during exposure without undue crowding.

Large-sized copper tubing was used to minimize the RF conductor losses and consequently the RF driving power requirements

of the source, as well as for reasons of mechanical strength.
The choice of a single -turn rather than a multiple -turn

inductor was made because of the following desirable characteristics : (a) a lower distributed capacitance;

(b) a higher resultant self -resonant frequency;

(c) a lower "unwanted" electric-field strength due to a lower reactive voltage drop; and

(d) less shielding of the "desired" electric field of

the associated parallel plates previously mentioned.

This shielding would in effect produce a distortion

of the spatial distribution of the "desired" electric

field which would increase with the number of turns

of the inductor.

All joints in the loop are high-temperature silver-brazed to

provide maximum electrical conductivity, mechanical strength,

and water tightness (for cooling to be discussed later)

2 . 2 The Electrical Characteristics of the Loop Inductor . The

inductor is driven, in effect, by four synchronous RF generators located symmetrically 90 degrees apart around the loop

periphery [1,2]. While these generators are in-phase with

respect to each other, diametrically-opposite generators

create electric fields which are spatially 180 degrees out

of phase and therefore cancel at the center of the inductor.

This provides for an order-of -magnitude (10:1) reduction in

the "unwanted" electric field that would otherwise exist in

the central portion of the loop if driven by only one generator as shown in Appendix A. The magnetic field remains unaffected since only those components of the electric field

that do not contribute to Curl E are being cancelled as will

be discussed later.

Instead of a single gap as found in the usual "shielded"

loop [3,4], this loop has four gaps (90 degrees apart) as
shown in Figure 1. The four driving voltages are fed-in

coaxially and appear across the respective gaps, in-phase

with respect to each other. The inner conductor of the coaxial drive is 3/4" (19.1 mm) diameter copper tubing with a 1/32 M (0.794 mm) wall thickness. With the electrical balance

or symmetry that must be maintained around the periphery of

the four-gap loop, four voltage nulls or zero -potential

points result, one midway between each of the four gaps.

These null points lie on two orthogonal image planes intersecting at the center of the loop. This permits the loop to

be both mounted mechanically and driven electrically from two

diametrically opposite null points as indicated in Figure 1.

Since each of the matching networks shown has a balanced (pushpull) output, each, in effect, provides two of the four driving

voltages, discussed above, in the proper phase relationship.

An exploded view of the four-gap loop inductor and the coaxial

inner conductors is shown in Figure 2. A view of the assembled

loop inductor and its associated matching networks is shown in

Figure 3.

The "unwanted" electric-field in the plane of the loop is

proportional to both the operating frequency and the distance

from the center, r, as shown by Equation (7) in Appendix A.

For a magnetic field strength of 50 amperes per meter, as specified by the sponsor, the electric field at a distance,

r, equal to one-quarter of the loop radius, varies from

approximately 130 volts per meter at 10 MHz to 400 volts per

meter at 30 MHz. This has been found to be approximately an

order of magnitude lower than would be the case if only a single driving generator were used as shown in Figures 14 and

16 in Appendix A. The magnetic-field generator is designed

to produce a field strength of up to 100 amperes per meter

with a transmitter RF power output of up to one kilowatt

which is the maximum RF power capability of the units supplied

by NBS.
0- MATCHING

NETWORK

NO. I T

REFLECTOMETER

TO RF

SOURCE

T

MATCHING

NETWORK

NO. 2 -

REFLECTOMETER

-LI

Figure 1. A Block Diagram of the RF Magnetic-Field Generator

Showing the Four-Gap Shielded-Loop Inductor and the Associated

Tunable Matching Networks for Use from 10 to 30 MHz. The Two

Orthogonal Image Planes are Shown Intersecting at the Center

of the Loop.
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2.3 The Loop- Inductor Matching Networks . The two balanced,

tunable networks provide for impedance matching between the

loop inductor and driving generator over the frequency range

10 to 30 MHz. The circuit diagram of one of the two identical

matching networks is shown in Figure 4, and comprises vacuum

variable capacitors C-l and C-2, and inductor L-l. A separate —tf

TO LOOP

*

TO RF

'SOURCE

C2

Figure 4. The Circuit Diagram of One of the Two Identical

Matching Networks Used with the Shielded Loop Inductor.

balun transformer is also provided, as shown, between the

300-ohm balanced input to each network and its 75 -ohm coaxial

transmission line. The coaxial input circuits of the two

matching networks are driven in parallel from a common source

as shown in Figure 1. A close-up view of one of the matching

networks is shown in Figure 5.

The loop inductor, and inductor L-l in Figure 4, are

tuned to resonance at the desired operating frequency by the

capacitors, C-l and C-2, each of which has a range of adjustment from 10 to 1000 pF . The balanced input impedance is

determined by the value selected for inductor, L-l, in a manner exactly analogous to that employed when using a tapped,

tuned tank circuit. When properly resonated the input impe-
Figure 5. A View of One of the Tunable Loop - Inductor Matchin

Networks Showing the Two Vacuum Capacitors and the Shielded

Balun Transformer.

10
dance, Z. , is purely resistive and is given to a good approximation [5] by the following expression (provided Q > 10)

:

X

2 Z in - -±, ohms, (1)

R

where X

T is the reactance of the inductor, L-l, at the resonant

frequency, and R represents half the total equivalent series

loss resistance of the loop circuit, including conductor, dielectric, and radiation losses. The latter is assumed negligible as will be discussed later. This means that essentially

all of the input RF driving power to the loop will be converted into heat and dissipated in the loop and associated

circuitry.

2.4 The Vacuum Variable Tuning Capacitors . The vacuum

variable capacitors used in the loop-inductor matching networks have the following advantages over conventional air

variable capacitors for this type of use:

(a) a larger absolute capacitance as well as a larger

range of capacitance adjustment (ratio of maximum

to minimum capacitance) for a given physical size;

(b; a higher RF voltage breakdown rating;

(c) lower RF losses and heating and therefore a higher

RF current rating. The RF current rating for a given frequency can be limited by either the RF

heating, or by RF voltage breakdown resulting from

the reactive voltage drop;

(d) a lower internal inductance, and therefore a higher

self -resonant frequency.

2.5 The Balun Transformer [6]. The toroidal balun transformer

used in each of the networks converts the 300-ohm balancednetwork input impedance to a 75-ohm unbalanced coaxial drive.

11
The balun has an RF power rating of 1 kilowatt continuous at

99 percent efficiency (.05 dB insertion loss). A view of the

balun is shown in Figure 6. The winding arrangement is shown

in Figures 7 and 8. Copper ribbon 0.013" (0.330 mm) thick by

0.100" (2.54 mm) wide is used which is the equivalent of No. 14

AWG (0.064" = 1.63 mm dia) copper wire in electrical conductivity in this frequency range and is much easier to wind

physically. The balun is wound on a ferrite toroidal core

approximately 2-1/2" (63.5 mm) O.D. x 1-1/2" (38.1 mm) I.D. x 1.0" (25.4 mm) thick which has an initial relative permeability of approximately 40 for frequencies up to at least

30 MHz. The purpose of the core is to increase the commonmode rejection at the lower frequencies. The transformer

has a measured common-mode rejection of approximately 30 dB

over the frequency range 10 to 30 MHz.

The winding directions are important and are as shown in

Figure 8. The common-mode currents, I , in the two transmission

lines magnetize the core in the directions shown by the arrows.

Since the magnetic flux produced by these currents in the two

lines is in the same direction in the core, this results in a maximum common -mode inductance. The balanced-mode currents in

the two sides of each transmission line are equal in magnitude

and opposite in phase. They therefore produce very little, if

any, magnitization of the core and hence no balanced-mode inductance. The core therefore has little, if any, effect on

the two transmission lines operating in the balanced mode and

serves only to attenuate the common -mode currents. As a result, there is little, if any, RF core loss in normal operation. The losses are essentially all in the copper windings,

which helps to explain the high operating efficiency.

2.6 The Water Cooling of the Inductor . The loop inductor is

designed to be water-cooled, since the measured RF driving

12
j- <*»* -^"^ >'-- - .^thara-,^--

Figure 6. A Bottom View of One of the 75 to 300 Ohm Toroidal

Balun Transformers Used in the Loop -Inductor Matching Networks.

13
Zo

Hi- 2Z,

Figure 7. A Basic Balun Circuit Comprising Two Balanced

Transmission Lines of Characteristic Impedance, Z . The

Lines are Connected in Parallel at One End and in Series at

the Other, Giving an Impedance Transformation Ratio of 4:1.

2Zr

Figure 8. The General Method of Winding Used in the Toroidal

Balun Transformer. The Basic Arrangement is the Same as that

of figure 7, Except that the Transmission Lines Have Been

aid on a Toroidal Core to Increase the Common-Mode Reiection

at tne Lower Frequencies.

14
power required to overcome circuit losses and produce a magnetic field of 50 amperes per meter is in the range from

approximately 175 watts to over 500 watts over the operating

frequency range 10 to 30 MHz. A 1/2" (12.7 mm) O.D. plastic

water hose connects to the nozzles which can be seen in

Figures 2 and 3 at the extreme ends of the center conductors. Only a minimal amount of water flow is required to keep

the loop at room temperature (approximately 1 quart (0.95

liter) per minute which can be obtained with ordinary tap

pressure is sufficient) . Water cooling is necessary for two

reasons: (a) it eliminates any possible effect of an increase

in the ambient air temperature on the primates during exposure;

and (b) it stabilizes the loop tuning which otherwise drifts

appreciably as the loop heats up. The water flows through

the interior of the loop center-conductor and so is essentially

not in contact with the electrical part of the circuit. This

is true except at points of entry and exit where the length

of the water column is sufficient to minimize any shunting

effect

.

3. THE RF ELECTRIC -FIELD GENERATOR

3 .1 The Physical Characteristics of the Parallel -Plate Strip

Line . The RF electric-field generator consists of a balanced

parallel -plate strip line fabricated from 1/16" (1.59 mm)

thick aluminum sheets approximately 30" (75 cm) wide and spaced

the same distance apart. The line is also approximately 30"

(75 cm) in length along the straight portion, and tapers both

in width and spacing over a distance of approximately 2 0"

(50 cm) on the driven end as shown in Figure 9. This permits

connecting the balanced RF feed line with a minimum of electrical discontinuity. Connection is made to the balanced

strip line by means of two large feed-through bushings using

15
TAPERED

FEED

E

o

in

I sLOWER PLATE

Figure 9. The General Arrangement of the EM Near-Field Synthes izer . The Loop Inductor is Located Midway between the

Balanced Parallel Plates Normally in the Neutral Plane, but

can be Tilted at Any Angle up to 60 Degrees.

steatite insulating cones. These pass through a metal plate

on one side of the shielded room in which the synthesizer is

located and into the tunable matching unit which is fastened

to the outside. A view of the strip-line is shown in Figure 10

3.2 The Parallel -Plate Matching Network . A balanced, tunable

network is provided for impedance matching between the parallel

plates and the driving generator. The network is housed in a large aluminum box 20" (50.8 cm) x 20" (50.8 cm) x 18" (45.7

cm) fastened to the outside center of the shielded room in

which the synthesizer is operated. Electrical connection is

made to the parallel plates by means of the two large feedthrough bushings previously mentioned. The size of the housing

is dictated by the size of the tuning coils employed in order

to obtain maximum Q and RF power-handling capability. A small

muffin fan is used to exhaust air from a vent in the top side

of the enclosure to facilitate cooling the inductors of the

16
Figure 10 . A View of the Electromagnetic Near-Field Synthesizer as Installed in the Shielded Room. The Parallel -Plate

Strip Line and the Four-Gap Loop with its Matching Networks

can be seen.

17
matching network. A matching air inlet is provided in the

bottom of the enclosure to ensure proper air circulation.

The circuit diagram of the matching network is shown in

Figure 11. The parallel plates are tuned to resonance at the

UPPER PLATE

LOWER PLATE

I—

C2 *£

TO RF

SOURCE

Figure 11 . The Circuit Diagram of the Parallel -Plate

Matching Network.

desired operating frequency by the balanced inductance comprised of L-l, L-2, and L-3. The two vacuum variable capacitors, C-l and C-2, facilitate the tuning. Each has a range of

adjustment from 6.5 to 50 pF. The input impedance is determined by the value selected for inductance L-2 in exactly the

same manner as previously described for the magnetic-field

generator, and can be determined from Equation (1), Section

2.3 . A toroidal balun transformer is used as shown in the circuit diagram to convert the 300-ohm balanced-network input

impedance to a 75 -ohm unbalanced coaxial drive. The transformer is identical to those used with the loop matching

networks

18
3.3 The Electrical Characteristics of the Parallel -Plate

Strip Line . The "unwanted" magnetic field produced by the

parallel plates was found to be negligibly small for the

plate size, frequency, and the magnitude of the electric

field, E, being generated. No special design was undertaken

to minimize this "unwanted" magnetic field as was done in the

case of the loop inductor. Although, it would have been

possible to have reduced the magnetic field in the central

portion of the parallel plates by a factor of at least three,

if necessary, by driving the plates simultaneously [7] at

both ends instead of at one end.

An RF driving power' of between approximately 150 and 300

watts is required over the frequency range of 10 to 30 MHz

to produce an electric field strength of 5000 volts per meter

between the plates as required by the sponsor. This is the

power required to supply the conductor and dielectric losses

of the parallel plates and associated circuitry, the radiation

losses again being negligible because of the mode of operation,

as in the case of the loop inductor. Because of the surface

area involved in the parallel plates, their losses are minimal.

Little temperature rise was observed, and no cooling was found

necessary. Most of the loss was found to be in inductors L-l,

L-2, and L-3 (Figure 11).

The electric-field generator is designed to produce a field-strength of up to 10,000 volts per meter with a transmitter RF power output of up to one kilowatt which is the

maximum RF power capability of the units supplied by NBS.

4. THE COMPLETE SYNTHESIZER UNIT

4.1 A General Description . In the completed EM near-field

synthesizer the single -turn loop inductor of the magnetic-field

generator is mounted midway between and parallel to the two

19
balanced plates of the electric-field generator as shown in

Figure 9. The normal position of the loop inductor is thus

in the neutral plane between the plates, although the loop

can be operated at any angle to the horizontal of up to 60

degrees. This provides the means for adjusting the relative

spatial orientation between the electric and magnetic field

vectors as specified by the sponsor. The electric and magnetic coupling between the two field systems has been

minimized because of the balanced symmetry employed in the

excitation of both the loop inductor and the parallel plates,

as well as their relative physical orientation. This was

confirmed during the acceptance tests at Brooks Air Force

Base in which no observable cross coupling could be detected

even with power inputs of up to one kilowatt to either or

both the field generators for any angular position of the

loop inductor up to 60 degrees from the horizontal.

4.2 The Effect of the Shielded Room . The near-field synthesizer is installed and operated inside of a doubly-shielded

copper room approximately 7 ft (2.13 m) wide x 10 ft (3.05 m)

long x 8 ft (2.44 m) high. The RF power required to drive

either the electric- or magnetic-field generator has been

found to be the result mainly of losses within the generators

and matching networks themselves (i.e. metallic and dielectric

losses in the inductors, capacitors, parallel plates, etc.).

Radiation losses are, for the most part, nonexistent because

of operation of the generators inside a copper- shielded room

having a shielding efficiency of at least 60 dB . Coupling

between each field generator and the interior walls of the

screened room has been found to be very loose and is essentially reactive. Very little resistance is coupled back into

the field generators from the room walls, since: (a) thi

copper walls of the room are low-loss; (b) the physical

ie

20
dimensions of the generators are small compared to the room

dimensions; and (c) the frequency of operation is well below

the lowest possible self -resonant frequency of the shielded

room operating as a low-loss rectangular cavity resonator.

For this room the lowest possible self -resonant frequency [8]

is approximately 80 MHz.

4.3 The RF Excitation of the Field Generators . The two field

generators are driven or excited as shown in the block diagram

of Figure 12. Separate, synchronized, transmitters are used,

each of which is composed of a 100-watt driver transmitter

followed by a one -kilowatt RF power amplifier as shown.

Synchronization is achieved by operating both 100-watt driver

transmitters from the same crystal oscillator. Separate frequencies can be easily used for each transmitter if for any

reason it is desired. Reflectometers are provided, as shown,

to facilitate tuning the two systems. It is found advantageous to first tune the loop-inductor and the parallel

plates so as to have the correct input impedance, using one

of the new RF vector- impedance meters supplied by NBS

.

4.4 The Phase -Shift -Networks . Two pi-type phase-shift networks [9] were designed and constructed at NBS to provide for

an adjustable time -phase difference between the electric and

magnetic fields generated, in addition to the adjustable

spatial orientation provided. These units are identical in

size and appearance to the two impedance matching networks

provided with the loop -inductor as shown in Figure 5. They

provide for phase shifts from 30 to 150 degrees each over the

frequency range 10 to 30 MHz. They can be used singly or

in tandem to provide a total phase shift of up to 300 degrees

if desired. The circuit diagram of the phase-shift networks

is shown in Figure 13.

21
1 cc OLU

UJr-

_JUJ ^UJDC o

<£> ^

-Z.CSL

<UI

S^

UJ _|CO OQ_jLU

z:lu<h < UJq;<

_ILl<-I < QlOL m o i , 1 /

MATCHING NETWORK

"1

ccUJ_

>^< — o

crz: Qi r J

Q_S~<o-z. cc£

Q_' ' i or OLD

Ull- 1 UJ hs o lo

,03o ^ J

C

•HT3

i-H rH

CD

U

•H

d

PL, I-H

r-i

CCJ CD

4-> cjCD

CJ 2 o6

CD

O

+->

u

CD

r-t rH O -H u u

cj

CD -H

4-> On

•-H CD O ^g

aj

m

?H o

Cti rH Quo

I-l pq

V) W) CD a u

0)d

b0| •H 3o

CO

CDo h a,

</) CD Ph

Ci CD CO 4->

22
Figure 13

.

The Circuit Diagram of the Pi-Type Phase-Shift

Networks Used with the EM Near-Field Synthesizer. These are

Similar to a Pi-Type Impedance -Matching Network Except they

are Inserted in an Already-Matched System to Produce an

Insertion Phase Shift Without Disturbing the Match.

4.5 The RF Exposure Capability . The electric-field generator

can produce a fairly pure (high-impedance) E field of up to

10,000 volts per meter over the frequency range 10 to 30 MHz.

The magnetic-field generator can produce a fairly pure (lowimpedance) H field of up to 100 amperes per meter over the

same frequency range. The two fields can be adjusted essentially independently in amplitude. As a result, the ratio of

the magnitude of the electric field to that of the magnetic

field can be adjusted to any value between roughly 40 and

4000. The electrical time -phase difference bet", ien E ana H

can be adjusted for angles up to 300 degrees, and relative

spatial orientation for angles up to 60 degrees in order to

simulate various near- field configurations. An RF driving

power of up to one kilowatt is required for each field system depending upon frequency and the field level used.

23
5. SUMMARY AND CONCLUSIONS

This publication describes the development, design, construction and testing of a prototype EM near-field synthesizer

conducted by the National Bureau of Standards for the USAF

School of Aerospace Medicine at Brooks Air Force Base. The

purpose of the contract was to provide a means of independently

generating high-level electric and magnetic near-fields in the

frequency range 10 to 30 MHz. These fields are to be used in

various ratios by the USAFSAM in their EM radiation exposure

program for determining the biological effects of hazard-level,

non-ionizing EM fields on human beings.

The synthesizer consists of: (a) a balanced parallelplate strip line to generate the "desired" electric field;

and (b) a single -turn quadruple -feed inductor placed parallel

to and midway between the plates to generate the "desired"

magnetic field. It is generally known from Maxwell's equations

that in addition to the "desired" E and H fields in (a) and (b)

respectively, the electric field in (a) is accompanied by an

"unwanted" magnetic field, and the magnetic field in (b) is

accompanied by an "unwanted" electric field.

In order to obtain essentially independent control over

the amplitudes of the "wanted" electric and magnetic fields

as produced above, it was necessary to reduce the "unwanted"

field components insofar as possible. It was shown how the

"unwanted" electric field associated with a single-turn loop

inductor was reduced in magnitude in the central portion

of the loop by as much as a factor of ten by increasing the

number of in-phase RF sources feeding the loop from one up to

four. It was also shown that the four-source loop was probably

the optimum from a practical standpoint, and the design of

such a loop was described in detail. It was found that the

"unwanted" magnetic field produced by the parallel-plate strip

24
line was negligibly small for the plate size, frequency, and

the magnitude of E used. Although the "unwanted" magnetic

field could have been reduced substantially by using a symmetrical dual feed to the plates had it been found necessary.

In order to make the amplitude adjustments of E and H

essentially independent it was also necessary to minimize the

electric and magnetic coupling between the two field systems.

This was accomplished basically by operating each generator

in a balanced (push-pull) mode and by physically placing each

generator in the neutral plane of the other. When the system

was installed at Brooks Air Force Base, there was no observable

cross-coupling between- the two field generators even when

operating at a full kilowatt^ of RF input power, or when the

loop was rotated out of its initial plane of symmetry by any

desired angle.

It was pointed out that the RF driving power for each

field generator is determined by the associated conductor-,

and dielectric-losses, the radiation losses having been minimized by enclosure in a copper-shielded room. The required

RF driving power has been found to be roughly proportional to

frequency. Water is circulated through the interior of the

loop inductor to maintain it at room temperature. This prevents any adverse effect of an increase in the ambient air

temperature on the primates during exposure, and stabilizes

the loop tuning which otherwise drifts appreciably as the

loop heats up. Special tunable networks and balun transformers

provide for impedance matching between the field generators

and their RF driving sources.

Under the conditions of use the single -turn inductor can

generate a fairly "pure" (low-impedance) H-field of up to 100

amperes per meter over the frequency range of 10 to 30 MHz.

The parallel -plate line can likewise generate a fairly "pure"

25
(high -impedance) E-field of up to 10,000 volts per meter over

the same frequency range. An RF driving power of up to one

kilowatt is required for each field system depending upon frequency and the field level used. The magnitudes of the

electric and magnetic fields can be adjusted essentially

in(j^-- " v over "

r ^de r; nge , as well as their relative

electrical time -phase, and relative spatial orientation in

order to simulate various, near-field configurations. Previous

esea ~h in RF biological hazards ha? been largely limited

the use of plane -wave fields for ma i g ^.iical studies.

This new device will allow researchers to investigate any

near-field effects that may occur at high-field levels.

6. ACKNOWLEDGMENT

The author wishes to acknowledge the support provided by

the USAF School of Aerospace Medicine at Brooks Air Force

Base, Texas, for the work described in this report. Appreciation is also expressed to Dr. James W. Frazer, Air Force

Project Officer, for his assistance and encouragement

throughout the project, and for his formulation of the

specifications on which the EM synthesizer is based.

The assistance of Drs . Howard E. Bussey and David M. Kerns

throughout tne project is also gratefully acknowledged, and in t particular their suggestion regarding the use of multiple

in-phase driving sources to reduce the "unwanted" electric

field associated with the magnetic - field generator.

26
7. APPENDIX A

THE ELECTRIC FIELD ASSOCIATED WITH A LOOP INDUCTOR

7.1 Introduction . First, the electric-field distribution

associated with a single-turn loop inductor will be evaluated

when the loop is fed from a single -source generator operating

at a frequency of 30 MHz. It will then be shown how this

electric field can be greatly reduced by the expediency of

using multiple in-phase driving sources or feeds distributed

symmetrically around the loop. A practical method of achieving the effects of the multiple feeds will also be shown.

The single-turn inductor used in these tests had the dimensions as previously given, namely, an average diameter, d,

,

of 21 5/8" (0.549 meter), formed from copper tubing having an

outside diameter, cL , of 1 5/8 M (0.0413 meter).

The low-frequency inductance of such a loop is given in

microhenries by the following expression [10] which neglects

distributed capacitance:

L = 0.2tt d

1 [2.303 log

1

8d.

L*2J

2 + V&], (2)

where d.. and d

? are in meters, and provided d,/d„ > 5. The

quantity y6 represents the internal inductance, which is

negligibly small for copper in this frequency range. For the

dimensions given, the low-frequency inductance, L, is

approximately 0.921 microhenry neglecting \i6. The apparent

inductance, L', increases with frequency due to partial

resonance resulting from: (a) the finite size of the loop;

(b) its distributed capacitance; and (c) the gap capacitance

(assumed to be 5 pF) . At a frequency of 30 MHz the apparent

inductance was found to be L' = 1.36L = 1.25 microhenries.

The apparent inductive reactance is therefore XJ = 2TrfL' = 236 ohms.

27
The maximum value of magnetic field strength, H, at the

center of the loop, specified by the sponsor is 50 amperes

per meter (A/m) . The relationship between H, and the driving

current, I, is given by [11]

H = —, A/m. (3)

1

7.2 The Single -Source Loop Inductor . An RF driving current

of approximately 25 amperes is required, as determined from

Equation (3), which results in a driving voltage of 5900 volts

at the loop terminals for an operating frequency of 30 MHz.

If the loop is driven by a balanced transmission line with a conductor spacing of 0.1 meter (4.0 inches) the electric field

strength in the plane of the transmission line will be of the

order of 30,000 volts per meter which will probably surprise

a lot of people. The electric field strength associated with

the single -source loop is proportional to the driving voltage

required to force the desired current around the loop. This

voltage is in turn essentially proportional to frequency

since V - I X-f (neglecting resistance). Therefore, the

electric field associated with a given loop will be greatest

at the highest operating frequency, namely 30 MHz in this

case

.

The measured electric-field distribution for the above

loop with a current flow of 25 amperes at a frequency of 30

MHz is shown in Figure 14. It can be seen that the electricfield strength is extremely high for this case and varies

from 2.3 to 16.0 kilovolts per meter for the positions shown

across the diameter of the loop.

28
Figure 14 . The measured electric-field distribution

associated with the single-turn loop inductor having

a single driving source at £ = 30 MHz. I

T = 25 amp.,

d, = 0.549 meter, V = 5900 volts.

1 £

7.3 A Double -Source Loop Inductor . The associated electricfield strength of the previous loop can be appreciably reduced (but not entirely eliminated) by the expediency of

using a multiplicity of driving sources properly phased. For

example, the driving voltage of the single source in the

previous case can be split into two equal parts and fed into

the loop at two diametrically opposite points around the

loop. The resulting two driving voltages must be in-phase

with respect to each other, so that the magnitude of the total

driving voltage remains unchanged. The two voltages create

electric fields which are spatially 180 degrees out of phase.

The resulting electric-field distribution is then as shown

in Figure 15.

What happens is that the electric fields of the two sources

are, in effect, superposed and subtract everywhere, resulting

in a complete cancellation at the center of the loop as shown.

The electric field is greatly reduced everywhere except near

the second source, V~ on the right, where it is somewhat

greater than before.

29
Figure 15 . The measured electric -field distribution

for the double -source loop inductor. V-. = V? = 2425

volts, f = 30 MHz.

7.4 A Quadruple -Source Loop Inductor . Now, if the total

driving voltage is divided into four equal parts, instead of

two, and injected every 90 degrees around the loop, such that

the four sources are in-phase with respect to each other, the

resulting electric-field distribution is as shown in Figure 16

V2

Figure 16 . The measured electric-field distribution

for the quadruple -source loop inductor. V, = V" 2 = V., = V 1150 volts, f = 30 MHz.

30
It can be seen that each time the number of sources is doubled

the electric field-strength in the vicinity of each source is

halved, since each of the driving voltages has been halved.

If Figure 16 is compared with Figure 14 it can be seen that

the electric - field strength near the center of the loop for

the quadruple -source case is only about 1/10 that for the

single -source case, so that a substantial reduction in E in

the central region of the loop has been achieved.

7 . 5 A Loop Inductor with a Continuously-Distributed Source . It soon becomes apparent that four sources is about the upper

practical limit, since the electric field in the central portion of the loop will not be appreciably further decreased if

the number of sources is further increased as will be shown.

The electric-field distribution corresponding to an infinite

number of driving sources distributed around the loop can be

derived theoretically from one of Maxwell's equations relating

E and H. For the sinusoidally time -varying , steady-state

case, this is [12]

V x E = -jgjuH. (4)

If both sides of equation (4) are integrated over the circular

area, A = irr 2

, and Stoke's theorem is applied, we obtain

2tt

j> E»d£ = -jcouHA. (5)

o

Due to the circular symmetry involved, E»d£ will be constant

around the circular path, £, of radius r and Equation (5) can

be written

27rrE = -jwyH(iTr 2

) (6)

or

E = -j -2—, V/m, (7)

31
where E is everywhere tangent to the circle of radius r. H is

normal to the plane of the loop and is essentially constant

over the area of the loop except near the loop conductor itself. It can be shown that there is only a second-order increase in H as r is increased from to somewhat less than r,

,

the radius of the loop [13]. The magnitude of E is therefore

essentially proportional to r, at least in the central portion

of the loop.

The theoretical electric- field distribution for this case

(f = 30 MHz) is shown in Figure 17, corresponding to a loop

current of 25 amperes (H = 50 amperes per meter) where the Elines are circles as shown. It can be seen that the magnitude

Figure 17 . Theoretical electric -field distribution

for a loop inductor with a continuously-distributed

source (H = 50 amp./m, f = 30 MHz, cL = 0.549 m) .

of E in the central portion of the loop with a continuously dis

tributed source is not significantly less than that of the

quadruple -source case shown in Figure 16. These values of

electric field strength represent the absolute theoretical

minimum possible for the value of H specified; i.e., 50 amperes per meter at 30 MHz.

32
8. APPENDIX B

METHODS USED TO SIMULATE THE MULTIPLE -SOURCES

DRIVING A SINGLE -TURN LOOP INDUCTOR

8.1 Introduction . This section will describe the procedures

used to simulate and measure the electric -field distributions

of the multiple -source loop inductor given in Appendix A. Advantage was taken of the fact that radial nodal planes exist

in these cases, making it necessary to actually provide only

a single driving source plus one or more image planes in order to establish the actual electric -field distribution within

a single 180 degree or 90 degree sector bounded by the plane/s

The portion of the loop used was formed of two coaxial conductors as will be shown. A gap was cut in the outer conductor

at the appropriate place. The RF voltage appearing across

this gap served as the loop driving-voltage and so could be

positioned as desired around the loop periphery. Sample

numerical calculations will be made at a frequency of 30 MHz

using conventional transmission-line equations [14]

.

8.2 The Shielded Half -Loop with a Gap at One End . Figure 18

shows such a 180-degree half-loop, (A), being operated above

a ground plane of "infinite" extent. The image of the halfloop is provided in the ground plane. When the full loop is

bisected by an image plane, its self impedance is halved.

Both the distributed capacitance and the gap capacitance are

doubled, but the relative frequency correction, given in

Appendix A, Section 7.1, remains unaltered. The equivalent

coaxial driving circuit, (B) , is terminated in an impedance,

Z R

= j 118 ohms, which is half the self -impedance of the full

loop (X' = 236 ohms) previously given (f = 30 MHz) . The line

length, £, is half the circumference of the full loop.

33
B.

Vc

-£/\ =.0863

cBS

Z R = jll8ft

Figure 18 . (A) The Shielded 180° Half-Loop with a Gap

at One End, situated above an Infinite Image Plane;

(B) The Equivalent Coaxial Driving Circuit for Calculating the Various Characteristics of the System using

Conventional Transmission Line Equations. f = 30 MHz,

Z - 41.5 ohms.

The electric - field distribution above the ground plane

will be identical to that of a full 360 -degree loop fed with a balanced voltage of double the value. Using a half-loop makes

it somewhat easier to set up and measure since only a singleended driving source and matching network are required. The

ground-plane used was an aluminum sheet 48" (122 cm) x 48"

(122 cm) x 1/8" (3.18 mm), and the dimensions of the loop

were as previously given. The characteristic impedance of

the coaxial driving line was 41.5 ohms. Teflon washers,

located approximately every 6" (15.2 cm), were used to support

the center conductor. These had a minimal effect on the

operation of the line owing to the relatively low frequency

and completely mismatched conditions used.

34
The driving-point impedance of the shielded loop was

measured at several frequencies over the range from 10 to 30

MHz to verify the equivalent coaxial driving circuit. In all

cases the agreement between the measured and calculated values

of input impedance was within ±5 percent. This means that it

is possible to accurately calculate the driving voltage, current, power and the actual gap voltage all for a loop current

of 25 amperes at the desired operating frequency using conventional transmission -line equations. Sample calculations of

some of these quantities will be made later. The measured

electric-field distribution for this case is shown in Figure

14 for the single -source case, except that the missing half

has been filled in.

8.3 The Shielded Half-Loop with a Gap at the Center . This

half-loop and its equivalent coaxial driving circuit are shown

in Figure 19 and, except for the position of the gap and its

consequences, has the same general appearance and dimensions

as the previous half loop shown in Figure 18. The gap

capacitance of 5 pF (Section 7.1) is now across only half of

the full loop, so that the frequency correction mentioned in

the previous section is now substantially less. This can be

seen from the fact that the terminating impedance is now

Z R

= j 97 ohms, rather than Z R

= j 118 ohms as for the previous case. In the present case with the gap at the center,

the image half-loop in the ground plane provides the effect

of a second gap, so that the resultant electric -field distribution above the ground plane is the same as for the

double -source loop, shown in Figure 15.

8.4 The Shielded Quarter-Loop with a Gap at the Center . The

arrangement of the quarter loop is shown in Figure 20. Here,

only one fourth of a full loop is used in conjunction with

two orthogonal image planes arranged as shown. The gap

capacitance is now across only one quarter of the full loop,

35
B. Vs

i/X = .043l

ZR

= J97&

Figure 19. (A) The Shielded 180° Half-Loop with a Gap at the

Center, Situated above an Infinite Image Plane; (B) The Equiva

lent Coaxial Driving Circuit for Calculating the Operating

Characteristics. f = 30 MHz Z = o 41.5 ohms.

A

B

vs<v

////////

Y—iVX = .02l6—*j

.) 0JZ^

Z R

= j46^

Figu r e 20 . (A) The Shielded 90° Quarter-Loop with a Gap at

the Center, Situated between Two Orthogonal Image Planes;

(B) The Equivalent Coaxial Driving Circuit for Calculating

the Operating Characteristics. f = 30 MHz, Z = 41.5 ohms.

36
making the frequency correction still less than in the previous

cases. The terminating impedance is now Z R

= j 46 ohms, compared to Z R

= j 59 ohms (half of Z R= j 118 ohms) . The missing

three quarter-loop segments together with their gaps are in

effect provided by the two image planes, so that the resultant

electric-field distribution within the 90-degree sector between the planes is the same as for the quadruple -source loop,

shown in Figure 16

.

It should be clear from Figures 18 to 20 that each time

the loop is bisected, the electrical lengths of both the loop

and the coaxial driving lines are halved. This raises the

self -resonant frequency of the system, and makes it possible

for the quadruple -source loop to operate at a much higher frequency than the single -source loop.

8.5 Calculation of the Loop Characteristics . The method used

to calculate the various voltage, current, and impedance relationships of the loop inductor and its coaxial driving circuit

will be outlined in this section using conventional

transmission-line equations. Sample numerical calculations

will be made for the case of the quarter loop at a frequency

of 30 MHz.

The relationship between the sending-end voltage, V-,

and the receiving-end current, I R

, of a lossless transmission

line of characteristic impedance, Z , when terminated at the

receiving end in an impedance, Z R

, is given by [14]

V

g

= I R

Z R

cos 3£ + j I R

Z Q

sin Ql, (8)

where £ = line length, meters; 3 = 2tt/X; and Z =41.5 ohms.

Substituting the parameters shown in Figure 20, in Equation (8)

with I R

= 25 amp., gives

V q

= j 25(46 cos 7.78° + 41.5 sin 7.78°) = j 1280 volts.

37
The gap voltage, V

R

, may be determined from the relationship,

V

R

= I R

Z R

= j 1150 volts,

which is the loop driving voltage used in Figure 16. The

terminating impedance, Z R

, is the impedance of the quarter

loop, as discussed in Section 8.4, where Z R

= jX' = j 46 ohms

as indicated in Figure 20.

The sending-end current, Ig, is given by [14]

I Z

I s

= I R

cos g£ + j -|-^ sin g£ (9)

o

I s

= 25(cos 7.78° - ^j sin 7.78°) = 21.02 amp.

Zg = Vg/Ig = j 1280/21.02 = j 60.89 ohms .

Zg represents the driving-point impedance at the sending-end

of the coaxial line which is really Z„ transformed by the

transmission line. This is the reactance which is tuned out

by each of the vacuum capacitors in Figure 4. In effect,

each capacitor tunes its own quarter of the full loop in

actual practice.

The RF driving power can also be determined, using the

complete, hyperbolic transmission-line equations [14]. This

would involve the accurate calculation of the RF resistance

of the loop inductor and the four coaxial feed lines [15] , and, of course, is more involved than the calculations made

above for the lossless case. A preliminary estimate of the

required RF driving power can also be obtained, when necessary,

from a scaled measurement made at a low-field level.
9. REFERENCES

[I] Jasik, H. (editor), Antenna Engineering Handbook,

pp. 6-1 to 6_-4 (McGraw-Hill Book Co., Inc., New York,

N.Y. , 1961) .

[2] Kraus , J.D., Antennas, pp. 155-172 (McGraw-Hill Book Co.,

Inc., New York, N.Y., 1950).

[3] Libby, L.L., Special Aspects of Balanced Shielded Loops,

Proc. IRE, 34, pp. 641-646 (Sept. 1946).

[4] Weeks, W.L., Antenna Engineering, pp. 56-61 (McGraw-Hill

Book Co., Inc., New York, N.Y., 1968).

[5] Terman , F.E., Electronic and Radio Engineering, pp. 50-57

(McGraw-Hill Book Co., Inc., New York, N.Y., 1955).

[6] Sabaroff, S., The Balun "Formalized," IEEE Trans. EMC,

Vol. EMC -16, No. 1, pp. 48-50, (Feb. 1974).

[7] White, D.R.J. , EMI Test Instrumentation and Systems,

Vol. 4, pp. 4.20-4.24 (Don White Consultants, Germantown,

Maryland, 1971)

.

[8] Ramo , S., Whinnery, J.R., and Van Duzer, T., Fields and

Waves in Communication Electronics, pp. 541-543 (John

Wiley $ Sons, Inc., New York, N.Y., 1965).

[9] Jasik, H., op. cit., pp. 2_0-27 to 20-28.

[10] Radio Instruments and Measurements, NBS Circular C-74,

p. 250, eq. 147 (Jan. 1937).

[II] Greene, F.M., The Near-Zone Magnetic Field of a Small

Circular-Loop Antenna, NBS J. of Res., 71C , No. 4,

p. 323, eq. 26 (Oct. -Dec. 1967).

[12] Ramo, S., Whinnery, J.R. , and Van Duzer, T., op. cit.,

p. 228-238.

[13] Greene, F.M., loc. cit.

[14] Guillemin, E.A., Communication Networks, Vol. II, p. 69

(John Wiley § Sons, Inc., New York, N.Y., 1935).

[15] Wheeler, H.A. , Formulas for the Skin Effect, Proc. IRE,

30, pp. 412-424 (Sept. 1942).

39
NBS-114A (REV. 7-73)

U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET

1. PUBLICATION OR REPORT NO.

NBS TN-652

2. Gov't Accession

No.

4. TITLE AND SUBTITLE

Development and Construction of an Electromagnetic

Near-Field Synthesizer.

3. Recipient's Accession No.

5. Publication Date May 1974

6. Performing Organization Code

7. AUTHOR(S)

Frank M. Greene

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

Boulder, Colorado 80302

10. Project/Task/Work Unit No. 2727433

11. Contract /Grant No. USAF

Memorandum of Agre

ment, dated 3 Jan. 1

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

USAF School of Aerospace Medicine, Brooks Air Force

Base, Texas 78235

13. Type of Report & Period Covered Final Report

1/3/72 to 6/30/73

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here.)

This publication describes work done by the National Bureau of Standards for the

USAF School of Aerospace Medicine at Brooks AF Base involving the development,

design, construction and testing of a prototype EM near-field synthesizer. The pur

pose of the contract was to provide a means of independently generating high-level

electric and magnetic near fields in the frequency range 10 to 30 MHz. These fields

are to be used in various ratios by the USAFSAM in their EM radiation exposure program for determining the biological effects of hazard-level, non-ionizing EM fields on luman beings

.

The synthesizer consists of a balanced, parallel-plate strip line to generate the

'desired" electric field, and a single-turn quadruple-feed inductor placed parallel to

and midway between the plates to generate the "desired" magnetic field. Methods

used to reduce the "unwanted" E- and H-field components associated with the above,

as well as the methods used to reduce the coupling between the two field systems are

discussed. The result is a synthesizer in which the electric- and magnetic -field components can be adjusted essentially independently over wide ranges of magnitude,

relative time-phase, and spatial orientation to simulate various near-field configurations . Previous research has been largely limited to the use of plane-wave fields for evaluating RF biological hazards. This new device will allow researchers to investigate

any near-field effects that may occur at high field levels.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons) Electromagnetic -field hazards; electromagnetic -field

synthesizer; electromagnetic radiation-exposure testing (non- ionizing); near fields;

RF biological hazards.

18. AVAILABILITY [^ Unlimited

~2 For Official Distribution. Do Not Release to NTIS

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