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PUBL/CATIONS

't NisrUnited States Department of Commerce

Technology Administration

National Institute of Standards and Technology

N/ST Technical Note 1510

QC

ilOO

U5753

NO. 1510

1999

Switched-Coupler Measurements for

High-Power RF Calibrations

J. Wayde Allen

NIST Technical Note 1510

Switched-Coupler Measurements for

High-Power RF Calibrations

J. Wayde Allen

Radio-Frequency Technology Division

Electronics and Electrical Engineering Laboratory

National Institute of Standards and Technology

325 Broadway

Boulder, Colorado 80303-3328

July 1999

^1^

U.S. DEPARTMENT OF COMMERCE, William M. Daley, Secretary

TECHNOLOGY ADMINISTRATION, Gary R. Bachula, Acting Under Secretary for Technology

NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY, Raymond G. Kammer, Director
National Institute of Standards and Technology Technical Note

Natl. Inst. Stand. Technol., Tech. Note 1510, 20 pages (July 1999)

CODEN:NTNOEF

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1999

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402-9325
Contents

1 Introduction 1

2 Theory of Operation 1

2.1 Cascaded-Coupler Technique (Perfectly Matched Condition) 1

2.2 Effect of Impedance Mismatch on the Cascaded-Coupler Equation 3

2.2.1 Effect of Coupler Removal on Mismatch 3

2.2.2 Mismatch Corrected Cascading Equation 4

3 System Design - 6

3.1 Hardware 6

3.2 Software 8

4 Uncertainty Analysis 8

4.1 Type A Uncertainty 9

4.2 Type B Uncertainty 9

4.2.1 Fractional Uncertainty in the Bolometric Power Readings 11

4.2.2 Uncertainty in the Determination of Mismatch 11

4.2.3 CoupUng Ratio Stability (Linearity) 12

5 Comparison Test Results and Summary 12

References 15

ni

Switched-Coupler Measurements for High-Power RF Calibrations

J. Wayde Allen

National Institute of Standards and Technology

325 Broadway

Boulder, Colorado 80303

This paper describes the design and operational theory behind the new high-power measurement system

recently developed at the National Institute of Standards and Technology (NIST). This new system

makes use of the established low power bolometer calibration service, and extends our measurement

capability up to 1 kW in the 2 MHz to 1000 MHz frequency range.

Keywords: Microcalorimeter; Microw^ave Calorimeter; Microwave Power Measurement; Microwave Power

Standard Waveguide; Microwave Power Standard

1 Introduction

In order to meet the demand for higher power caUbrations, a new high-power measurement system had to be developed. The 'Cascaded-Coupler' technique, originally discussed by Kenneth Bramall in 1971 [1], was

chosen for its ability to achieve an acceptable uncertainty at minimum cost. Specifically, it allows us to

transfer the calibration of a bolometer mount measured on the existing microcalorimeter system to 1 kW

with measurement uncertainties on the order of 2 %.

2 Theory of Operation

2.1 Cascaded-Coupler Technique (Perfectly Matched Condition)

In the cascaded coupler technique, the calibration of a low-power (10 mW) bolometric power sensor is sequentially transfered to similar sensors on the sidearms of a coupler chain as shown in Figure 1. The 10 dB

coupler increment was chosen so that the bolometric sensors operate in the range from 1 mW to 10 mW.

This is the range over which these sensors in combination with the NIST Type IV power meter have the

lowest measurement uncertainty [2]. © © © n

-^N\t-^ -W\^^ -vW^^ ^AAA—

-^A^A-^

u^

40 dB K5)

Tg

I— Load

Figure 1: Series directional coupler chain.
Calibration of the coupler chain begins by connecting the calibrated sensor to the output port of the coupler

chain and adjusting the power until this sensor reads 10 mW. The sensor on the sidearm of the 10 dB coupler

should then read approximately 1 mW due to the 10 dB coupling ratio. It is now possible to determine

the ratio between the reading on the calibrated sensor and that of the 10 dB sidearm. This calibrates the

coupling ratio for the 10 dB coupler, and to the extent that the system is linear, the ratio will remain

constajit. For now, let us assume perfectly matched and linear conditions so that the effects of impedance

mismatch and nonlinearity can be ignored.

Replacing the calibrated sensor with a matched load gives us

Pi Plx' ^'

where:

Pg is the power delivered to the calibrated mount (approximately 10 mW),

Pi is the power read from the 10 dB sidearm (approximately 1 mW),

P/i is the power dehvered to the load, and

Pix is the 10 dB sidearm power with Pn dehvered to the load.

Solving eq. (1) for Pn gives

Pn = ^Ps. (2)

If the power is now increased until the 10 dB coupler sidearm power reads approximately 10 mW, the power

at the sidearm of the 20 dB coupler will be close to 1 mW. Forming a new set of ratios between the detectors on the 10 dB and 20 dB coupler sidearms results in — = —

,

(3)

where:

Pii is the load power referenced to the 10 dB sidearm,

P? is the power reading on the 20 dB sidearm with 10 mW on the 10 dB sidearm,

P/2 is the load power referenced to the 20 dB coupler sidearm, and

P2x is the 20 dB sidearm power with Pio delivered to the load.

Substituting eq. (2) in place of Pn and solving for P;2 gives

Pl2 = ^^Ps- (4)

At this point, the 10 dB coupler is no longer needed, and must be physically removed from the coupler chain

before the power is further increased. Increasing the power without removing the coupler will overload the

power detector on the 10 dB coupler sidearm.

Increasing the power, forming the ratio of powers, and physically removing the lowest power coupler stage can be continued to extend the power calibration as far a^ needed. For the coupler chain diagrammed in Figure 1, the power equation for the final 50 dB coupler is

p _ -Psi Pjx Pzx P2x P\x p ,cs

' ~ Ps Pa Pz Pi Pi " ^ ^
n.)

P3

b, a, aj b2

Figure 2: Single coupler circuit.

2.2 Effect of Impedance Mismatch on the Cascaded-Coupler" Equation

In the preceding discussion it was cissumed that all of the impedances are matched. In general they are

not, and it is necessary to include a mismatch correction in the cascading equation. Because the mismatch

correction depends on the equivalent generator reflection coefficient (Tg) of the coupler chain, it is important

to understand how removing couplers from the chain affects the measurement.

2.2.1 Effect of Coupler Removal on Mismatch

To understand the effect of coupler removal on the mismatch correction, the definition of Tg must be

considered. This definition has been documented in several places by Glenn Engen [3] [4, pp. 40-44], but can be summarized as follows. Consider the circuit shown in Figure 2. If we take a look at the scattering parameter representation of the

emergent wave amplitude 62 we get [4, eq. (6.5), p. 43]

h = hi-^ \-TdiS23

^31

521533 , 52l532

-)) + a2(522

531 531

(6)

where F^ = a^/b^. Since we are not changing the coupler or the sensor, both the S-parameters and F^ axe

constant, and we can simplify the scattering parameter representation by writing it in the form

62 — b^Scoupler + a2Fp,

where F^ is the 'equivalent' reflection coefficient given by [4, eq. (6.6), p. 44]

521 532

522 —

'31 and

^coupler

S2I , T^ fc S21S33. — h i d(.J23 ^

•JSl '^31

(7)

(8)

(9)

In other words, the scattering parameter representation of the coupler can be replaced with the microwave

equivalent of 'Thevenin's theorem' [4, pp. 17-20] as in Figure 3. Of particular interest is that Tg depends only on the internal characteristics of the directional coupler, not on anything that precedes the coupler. More importantly, the ratio of the power delivered to the coupler's

sidearm and the power delivered to its load are not affected by anything preceding the coupler.

Now note that during the initial calibration only the 10 dB coupler is used. The other couplers contribute

only their insertion loss, so the entire chain can be thought of a^ one large 10 dB directional coupler as shown

in Figure 4.
u^

bj - bj Scoupier + 82 r"g

Figure 3: Equivalent generator representation of a directional coupler.

©

K<5)

Load

Figure 4: Equivalent circuit for first stage calibration.

Here the only F^ of interest is the one measured for the entire coupler chain referred to the 10 dB coupler

sidearm and it can be used along with the values of F^ and F; to compute the impedance mismatch between

both Ps and the test load.

In the next step, transferring the calibration from the 10 dB coupler to the 20 dB coupler sidearm, the

power is increased until we can read the power on the 20 dB sidearm. Nothing else is changed, but we now know how the power delivered to the load changes with respect to both the 10 dB and 20 dB sidearm

powers. At this point, since the relationship between the 20 dB coupler sidearm power and the load power

has been determined, the 10 dB coupler can be removed without affecting the calibration. The transfer to subsequent couplers continues in the same way; as long as the test load is not changed, only the initial mismatch conditions need to be considered.

2.2.2 Mismatch Corrected Cascading Equation

Since we are not able to construct a measurement system using perfect power sensors and perfect impedance

matching between the components, eq. (2) must be modified to describe the more general condition. This is accomplished by noting that the level indicated by any of the power meters will only be the power delivered

to the meter multiplied by the bolometer's effective efficiency (77), so

(10)

(11)

(12)

where:

-Psm^i^T- is the DC substituted power reading on the meter connected to bolometer Ps,

Ms is the mismatch between the coupler chain and the bolometer mount Pg,

T]s is the effective efficiency of bolometer P,,

Ps is the net power delivered to bolometer Pg ,

Pimttcr the power indicated by the sidearm meter Pi with P3 attached,

Ps„\eter = MsTJsPs,

Px^.... = Ml 771 Pi,

^lx„^eter = M1771P1X,
Ml the mismatch between the coupler sidearm and the bolometer Pi

,

Tji the effective efficiency of meter Pi

,

Pi is the net power delivered to meter Pi

,

Pixmet^r- ^he powBr indicated on meter Pi after attaching the load and increasing the system power, and

Pix is the net power delivered to meter Pi after attaching the the load and increasing the system power.

Similarly, the power delivered to the load is given by

Pu.u.....=MiPi, (13)

where:

Pi is the net power delivered to the load, and

Ml is the mismatch between the coupler chain and the attached load.

If eq.s (10), (11), (12), and (13) are solved for the net delivered powers we get

^^ = %^' (14)

Pi = %^, (15)

p

Pi = ^''mT - (17)

These equations can then be substituted into eq. (2) to get

'"—-~pz:z^s~^- ^'^^

Ml and Ms are defined [4, eq. (3.16), pp. 20-22] as

Ml = ^^"i:" 'Xr. ,r"

' (19)

and

Ms = ^^

';"' ^^;, X''

(20)

|i i si g\ Substituting these into eq. (18), and generalizing for multiple coupler stages, results in the corrected cascaded

coupler equation

(1--|r/p)(i-|r,p)

|i-r^r,p

(1- |rsp)(i-|r,p)

'dclivcTcd PnXrr.^,er -^21^^,,, -Pll„,„r Psm^t^. I - \T

I

" "" ^

Primete. P2m.t.r- A^^ter Vs 1 " \^ s\-

1 - r^r.

1 - r,r.

(21)
Instrument Control Computer

DVM #1 DVM #2

Scanner

Type IV Power Meter Type IV Power Meter Type IV Power Meter Type IV Power Meter

U^

Type IV Power Meter Type IV Power Meter

Coupler Chain

-<3eLoad -•

Test Plane

Figure 5: Block diagram of measurement system.

3 System Design

3.1 Hardware

The previous discussion shows that the hardware requirements for the construction of a cascaded-coupler

system are relatively modest. Only the couplers required to create the chain of couplers, power meters for the

coupler sidearms, and an RF signal source capable of delivering the required power are needed. The coupler

chain itself can be manually connected and the couplers removed as the measurement proceeds. This design

works well for the occasional measurement, but is less attractive when a regular measurement program is anticipated. With this in mind, the system developed at NIST is designed to perform these functions under

computer control. Automating the measurement of the bolometer power levels is quite straight forward, and is achieved through

the use of commercial digital voltmeters (DVMs), NIST Type IV power meters [2], and a commercial switch

control unit or scanner. Figure 5 shows a block diagram of the system.

Two DVMs have been used to minimize the time difference between the power meter readings. Additionally,

two Type IV power meters are connected to each bolometer mount. One of the power meters and the

compensation element serve as a reference voltage generator (RVG). The other meter is connected to the

RF measurement thermistors. This helps correct for thermal drift in the bolometer mount, and keeps the

output voltage from the combined power meters in the DVMs most accurate operating range [5] [2]. Figure 6

shows the wiring diagram for each bolometer. With this arrangement, we can read the voltage directly from

the Type IV meter connected to the RF measurement bead by opening switch S2 and closing Si, or to read

the voltage difference resulting from the series connection of both meters by opening SI and closing S2. Converting the DVM readings to RF power follows the procedure described by Fred Clague in NISTIR 5016 [5]. Switch SI is initially closed and the voltage read with no RF power applied to get the initial zero power

reading Vn at time ti. Next, switch Si is opened and switch S2 is closed, giving a new off power voltage

Vixi with the series combination of the Type IV power meters at time ^2- The RF power is then turned on, and with switch S2 still closed the voltage V2x is meeisured at time ^3. Then after the RF power heis been

turned off, the voltage of the series power meter combination Vixf at time ^4 is measured. Finally, switch

S2 is opened and switch SI closed. The final off-power voltage Vi/ is then measured at time t^. Combining
'SW-^ ^HW^'

RF Measurement Compensation

Element Element

Bolometer

Figure 6: Power measurement circuit.

these measurements so that

and

V^i =Vu+[ ?M^ ) iV,f - Vu)

AV = V2, Vl^^ +

ts-h

tz-t2

(vix/ - Vi,o

f4 — ^2 . provides the values needed for substitution into the Type IV power equation [5] P =

KbRo

(2Fi - AF)AF,

(22)

(23)

(24)

where Rq is the operating resistance of the mount, and Kb is the mount cahbration factor. Computer control of the cascaded-coupler chain is achieved using a matrix of RF switches connecting the

couplers together as shown in Figure 7. This allows the computer to set the switches so that all the couplers

are connected at the start of the measurement cycle and to remove the appropriate couplers as the power

level is increased. The added insertion loss and power-handling capacity of the switches does, however, put

a limit on the power that can be switched.

Typical insertion loss for high-power coaxial switches is on the order of 0.2 dB, so the inclusion of the eight

switches shown in Figure 7 adds about 1.6 dB of loss to the coupler chain. Ignoring the insertion loss of the

couplers themselves means that the RF source would need to supply nearly 1.5 kW in order to overcome the

loss in the switching matrix. This exceeds the 1.1 kW power rating on the switches used, and the maximum

power capabiUty of our signal source. Simply excluding the final 50 dB coupler from the switching matrix

minimizes this problem. With this arrangement, the operator manually connects the 50 dB coupler and

calibrated bolometer Pg to the output of the switched coupler matrix before starting the measurement. The

computer initializes the coupler matrix by connecting all of the couplers in series and measures the power

ratio between F, and Pi £is described in Section 2. The operator then replaces Pg with the RF load, and the
Figure 7: Switched coupler chain.

computer automatically transfers the power calibration down the chain. This sets up the calibration for the

final 50 dB coupler, and is sufficient for power measurements up to 100 W. In this mode, the generator only

needs to supply around 150 W to overcome the switch losses. To measure powers higher than 100 W, the

operator simply disconnects the 50 dB coupler from the switched coupler matrix and reconnects it directly to the RF generator. The power can then be increased until 1000 W appears at the output of the 50 dB

coupler, and assuming that the coupler has about 0.2 dB of insertion loss, the generator only needs to supply

about 1050 W.

3.2 Software

The software for the measurement system was designed on the premise that it is easier to test, troubleshoot,

and maintain several small programs rather than one very large program. This allows the code development

to be split among the people working on the project and has the added benefit of the separate programs

performing consistency checks on each other's data. For this project, the software was split into three main

application groups: instrument control, data analysis, and report generation.

The instrument control program is the interface to the actual measurement hardware. It defines the measurement process and collects the data for final analysis. Only a minimal amount of computation such as the conversion of the Type IV DVM readings to RF power are done at this level. Data collected by the

instrument control program are made available for subsequent processing as a pure ASCII file. Data are analyzed by several Perl programs. One program formats these data into a more human-readable

form (bramview) and another computes the Type B uncertainty (bramerr). Another program formats the

output from the bramerr program into an ASCII grid suitable for importing into almost any spreadsheet

program. This allows the results from a series of several measurement cycles to be collected together into a

single spreadsheet, summarized, and the Type A uncertainty computed. The spreadsheet also facilitates the

creation of a table of results that can be imported into a word processing program for the generation of the

final report.

Ultimately, as we gain more experience with the idiosyncrasies of the system, the final data analysis and report

generation functions may be automated. With the highly modular design of the systems data structure,

adding this functionality or changing the analysis does not require a major rewrite of any large pieces of

code with the subsequent danger of breaking the existing functionality.

4 Uncertainty Analysis

Following the conventions described in NIST Technical Note 1297 [6], the system uncertainty analysis is composed of three fundamental parts: Type A (Ua), Type B (Ub), and the expanded uncertainty (U).

Type A uncertainties are those based on any valid statistical method for data analysis. Type B uncertainties
are those determined by any other method. The expanded uncertainty is

U = 2y/[/2 + f/2. (25)

The following sections describe how the Type A and Type B components were determined for the cascaded

coupler system.

4.1 Type A Uncertainty

The cascaded-coupler system is normally used to compare the power indicated by a high-power detector

{Pdut) to the calibrated power delivered to the detector by the coupler chain {Pi). The ratio of these two

powers

K = -^^

(26)

' 'deliver-ed

is the calibration factor for the meter and can be computed at any arbitrary power during the calibration

of the coupler chain. Since the calibration factor K is the value reported to the customer, this is the term

that must ultimately be assigned an uncertainty.

In this case, we want to determine the Type A uncertainty by computing the standard deviation of the mean

values of K determined at each power over at least three full calibrations of the cascaded coupler chain, or

y/n V n - 1

where:

Ua is the sample standard deviation of the calibration factor K,

Ki is the calibration factor for any single measurement,

Kavg is the average calibration factor, and n is the number of measurements made.

One difficulty with this approach is that it assumes that between any two calibrations the measurement

system can be reset to any given power. In practice, the power can only be reset to within several tens of watts. Fortunately, over this range the variation in K doesn't seem to be highly correlated with this variation

in power. This indicates that a more complex multivariate statistical approach is not really warranted.

Instead, the fluctuation in power is simply ignored, and the average of the measured powers is reported.

This effectively centers the reported value at the center of the cluster of data, and allows us to compute the

random variation or Type A uncertainty of the calibration factor K as given by eq. (27). Figure 8 shows

how the reported value of K and its Type A uncertainty compare for a typical set of data.

4.2 Type B Uncertainty

Several modifications to eq. (21) simplify the Type B uncertainty analysis. First, we can group the reflection

coefficient terms together to define a general mismatch term

M

and, second, we can define

1 - r.r.

1 - r,r,

(28)

p; =

^'"""^

. (29)
1.0038

1.0036 -

1.0018 105 Power (watts) 115

Figure 8: The reported calibration factor K and its Type A uncertainty for a typical raw data set.

This allows us to write the solution for the first coupler stage a^

(30)

Next, taking the partial derivatives for each of the terms in eq. (30) and computing the fractional change in

^'id.iiv,=r=d results in the expression

^I'idclivcrtd * ll^

APi , AP' AM

-Plm...r- M

(31)

An additional uncertainty term can be found by noting that eq. (21) assumes the directional couplers used

in the system are linear. That means that the coupling ratio

Ci- P'

should be constant. Equation (32) can be substituted into eq. (30) to get

P/1j I J ^= Pjt . C\M,

(32)

(33)

which indicates that the fractional change in Pad^/.^er.d due to a change in the coupling ratio is ACi/Ci.

Adding this coupler nonlinearity term to eq. (31) gives us an expression for the fractional uncertainty due

to the first coupler stage

AP,i, , , APix , APi , AP; AM ACi

PlUr '•aeiiveread ~ Plx^•''metcr , Pi>- meter , P'3 M Ci*

The solution for the second stage equation can be written in terms of eq. (30) a^

I'^dclivtrcd '' 'Idelivered •

(34)

(35)

10
The approach used to derive eq. (31) is then applied to eq. (35) to obtain

-B = -B + -5 + -7^ + -15, ' (36)

where C2 is fractional change in the coupling ratio for the second directional coupler. Equation (36) can

further be generalized so that the total uncertainty of any stage following the first one can be written as

p ~ p p r p

' ^ ' ^IndeliitT-ed -T nXm,(er ' "meter '-^n '("" 1 ) delivered

Referring back to eq. (26) we note that the calibration factor K only depends on the two values Pdut and

Pidei.ver-ed- SlncB for auy given measurement, Pdut is just the power read off the customer's device, the

uncertainty Ub only depends on the uncertainty in P/d,,ive--ed' '^^ AD

Ub =

'"'"""^'^^^ (38)

nnjel.vered

We assume a rectangular distribution for each of the terms in UbAll that remains is to find accurate estimates for each of the uncertainty terms:

• The fractional uncertainty in the bolometric power readings (APj/Pj and APn/Pn),

• The fractional uncertainty in the mismatch term (AM/M), and

• The coupling ratio stability or nonlinearity term (ACn/C'n)-

4.2.1 Fractional Uncertainty in the Bolometric Power Readings

The calibrated power P^' is determined from the DC substituted power indicated by the Type IV power

meter and the calibrated value of the mount's effective efficiency {rjg) as P ± AP

Vs ± A?7s

Consequently, the uncertainty in the value of P,' is

AP; = y/APlZ^^A^,. (40)

The value Arjs comes from the calibration of the mount Pj. The value for AP^^,,^^ is computed from the

uncertainties in the Type IV power meter and the voltmeter used to read the bias voltages. This computation

has been coded directly into the software used to compute the DC substituted power since the raw power

meter voltages are most readily accessible at this level. The computation of the DC-substituted power

uncertainty when using the NIST Type IV power meter is covered in references [2] and [7]. For the power measurements made on the coupler sidearms, a correction for effective efficiency is not needed

since these mounts are calibrated 'in place' against the mount P^. Only the measured DC substituted power

uncertainty need be considered.

4.2.2 Uncertainty in the Determination of Mismatch

Measurement of the reflection coefficients F/, F^, and F^ are made using either a vector network analyzer

(VNA) or six-port network analyzer. The resulting uncertainties are on the order of 0.005 in the veal and imaginary components. The uncertainty analysis software (bramerr) computes the mismatch from the

measured reflection coefficients Ti, Fg, and F,,, and the 64 possible variations in the mismatch due to the

addition or subtraction of 0.005 from the real and imaginary components. The mcLximum difference between

M and one of the 64 mismatch variants is used as the mismatch uncertainty AM

.

11
High Power

Source

Low Power

Source -W^ 1 KW Load

Attenuator

(a)

Figure 9: Test setup for determining coupler non-linearity

4.2.3 Coupling Ratio Stability (Linearity)

Equation (21) assumes that the system is linear and that the measured power ratios are constant. Unfortunately the world is not perfect and we need to allow for the possibility that the ratios depend on power.

A direct measurement of the power coupling ratio is not possible because it would exceed the dynamic

range of the available power sensors and is the reason for building the cascaded coupler system in the first place. However, assuming that changes in the coupling ratio are most likely caused by thermal expansion

in the coupler, we can design an experiment to get a reasonable estimate for the uncertainty due to coupler

nonlinearity.

The setup used for determining coupler nonlinearity is shown in Figure 9. The switches can be set so that a low power source can deliver a signal that can be sampled by both power detectors P3 and P2. The

attenuation factor {a) is chosen so that the detectors P3 and P2 are operating within their normal operating

range and is assumed to be constant. These powers can then be used to compute the coupling ratio

C =

P:iQ

(41)

Setting the switches so that the output of the coupler is connected to the 1 kW load and applying a high

power signal heats the coupler with the RF in the same manner in which it is used in the coupler chain. The

powers P3 and P2 can be remeasured on the 'hot' coupler by turning off the 'heating' power and immediately

resetting the switches to the low power circuit. The fractional uncertainty due to coupler nonlinearity is then

AC

c Ccold

(42)

where Ccoid is the coupling ratio measured before applying the heating power, and Chot is the resulting

measurement afterwards.

Tests done on water-cooled couplers gave us values for the fractional uncertainty ranging from 0.00008 to about 0.0005. Since this is such a small value, I decided that creating a separate uncertainty for each coupler

wcis not worth the extra effort. The maximum value of 0.0005 was chosen as being a reasonable estimate of

the fractional uncertainty due to nonlinearity in each coupler stage.

5 Comparison Test Results and Summary

To ensure continuity between the new switched-coupler measurement system and the older, low-frequency

Bramall system [8], comparison measurements were done on a feed-through wattmeter. Figure 10 shows how

these two systems compare at 30 MHz. This is the only point where we have measurement data up to 1000

W on the low frequency system. The figure shows that there is good agreement between both measurement

systems.

Table 1 shows a representative set of data taken with the new measurement system. The AP/P uncertainty

is the sum of the power ratio uncertainties given in eq. (37).

12
Table 1 : Table of typical measurement data.

Frequency Power

^ ^ ^ K Ua Ub U

(MHz) (W) (%) (%) (%) (%) (%) (%) (%)

550 0.572 0.05 0.24 0.10 0.992 0.71 0.40 1.63

550 13.525 0.08 0.24 0.15 1.003 0.83 0.47 1.91 550 93.424 0.10 0.24 0.20 1.019 0.79 0.54 1.92 550 993.854 0.12 0.24 0.25 1.025 0.82 0.61 2.04

1.05

1.04

1.03 g

I

1.02

<oC

mO

1

0.99

0.98

0.1

Historical Data >- Switched-Coupler System i—x10 100 Power (watts) 1000 10000

Figure 10: Comparison of historical calibration data with the new switched-coupler system at 30 MHz.

13
As shown here, typical uncertainties He between 1 and 2 %. We do see some degradation of the system

uncertainty at frequencies around 850 MHz and higher caused by a decrease in repeatabihty between multiple

calibrations of the coupler chain. However, this could be due to increased amplifier noise. Work is continuing

in an attempt to reduce the random uncertainty in the higher frequency bands.

14
References

[1] Bramall, K. E. Accurate microwave high power measurements using a ceiscaded coupler method. Journal

of Research of the National Bureau of Standards, 75C(3 and 4):185-192; July-December 1971.

[2] Larsen, N. T. A new self-balancing DC-substitution RF power meter. IEEE Transactions on Instrumentation and Measurement, IM-25, 4:28-32; December 1976.

[3] Engen, G. F. Amplitude stabilization of a microwave signal source. IRE Transactions on Microwave

Theory and Techniques, MTT-6, pp. 202-206; April 1958.

[4] Engen, G. F. Microwave Circuit Theory and Foundations of Microwave Metrology. IEEE Electrical

Measurement Series 9. Peter Peregrinus Ltd., London, United Kingdom; 1992.

[5] Clague, F. NIST model PM2 power measurement system for 1 mW at 1 GHz. NISTIR 5016, National

Institute of Standards and Technology; December 1993.

[6] Taylor, B. N.; Kuyatt, C. E. Guidelines for evaluating and expressing the uncertainty of NIST measure- ment results. Technical Note 1297, National Institute of Standards and Technology; 1994.

[7] Larsen, N. T. NBS Type IV RF power meter operation and maintenance. NBSIR 77-866, U.S. National

Bureau of Standards; October 1977.

[8] Rebuldela, G.; Jargon, J. A. High power CW wattmeter calibration at NIST. Journal of Research of the National Institute of Standards and Technology, 97(6):673-687; November-December 1992.

15

*U.S. GPO 1999-0-773-022/40 144

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