NIST Measurement Services:

NEW NIST PUBLICATION

June 12, 198?

AC-DC Difference Calibrations

U.S. Department of Commerce

National Institute of Standards and Technology
Center for Basic Standards

The Center for Basic Standards develops and maintains the scientific competencies and laboratory facilities nec- essary to preserve and continue to refine the base physical quantities upon which the Nation's measurement

system is constructed. The Center is responsible for the standards of voltage, current, resistance and

impedance, temperature, vacuum, leak rate, pressure, mass, length, time and frequency. The Center collaborates closely with the standardizing laboratories of other countries to ensure that measures used in international

trade are fully compatible. The Center also performs research at the frontiers of physics in order to improve

measurement capability and quantitative understanding of basic physical processes that underlie measurement

science.

Electricity Division

Maintains and improves the national standards of electrical measurements; conducts experiments to realize the

electrical units in terms of the mechanical units; develops stable electrical standards and improved methods for the dissemination of the units of electrical measurements; determines fundamental physical constants related to

electrical units; develops instrumentation for precise electrical measurements directly referenced to the national

standards; implements techniques for their use and puts them into practical applications; assists in applying scientific and technological development to measurement problems, especially in other areas of NIST research.

Temperature and Pressure Division

Maintains and improves the national standards of temperature and pressure; conducts research towards the development of new concepts for these standards; calibrates practical standards for the U.S. scientific and technical community in terms of the primary standards; develops methods and devices to assist user groups in the

assessment and enhancement of the accuracy of their measurements of temperature and pressure; prepares

and promulgates evaluations and descriptions of temperature and pressure measurement processes; coordi- nates these standards and measurement methods nationally and internationally; and determines the accuracy of the national standards of temperature and pressure with respect to fundamental thermodynamic relations.

Length and IVIass Division

Develops and uses the competence necessary to implement a more accurate measurement system for length

and mass; applies research, utilizing the most advanced technology available, to the improvement of our present

standards and devises new standards and new methods for their transfer. Time and Frequency Division

Maintains, develops, and improves the national standards for time (which are coordinated with the United States

Naval Observatory) and frequency, and the time scales based on these standards; carries out research in areas

of importance to the further fundamental improvement of frequency standards and their applications focusing on

microwave and laser devices, atomic and molecular resonances, and the measurement of ifundamental physical

phenomena and constants; adapts time and frequency standard devices and concepts to special scientific and

technological demands; develops time and frequency measurement methods in the radio frequency, microwave,

infrared, and visible radiation regions; coordinates the national time and frequency standards, time scales, and

measurement methods nationally and internationally in conjunction with the United States Naval Observatory;

operates time arKl frequency dissemination services, such as radio stations and broadcasts, for the purpose of

traceability to the national standards of time and frequency.

Quantum Phiysics Divisioi^

Engages in research in atomic and molecular physics at the forefront of the field, and performs basic, highly

accurate measurements and theoretical analyses that help establish a reliable foundation for scientific and technological measurement and data.

Quantum IVIetrology Division

Engages in forefront research in quantum metrology; contributes to new determinations of the fundamental

physical constants, and the extension and refinement of the electromagnetic scale and devises, where possible,

tests of basic symmetries arxl invariances.
NIST MEASUREMENT SERVICES:

AC-DC Difference Calibrations

J. R. Kinard, J. R. Hastings, T. E. Lipe, and C. B. Childers

Center for Basic Standards

National Measurement Laboratory

National Institute of Standards and Technology

Gaithersburg, MD 20899

May 1989

U.S. DEPARTMENT OF COMMERCE, Robert A. Mosbacher, Secretary

NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY

Raymond G. Kammer, Acting Director

NOTE: As of 23 August 1988, the National Bureau of Standards (NBS) became the National Institute of Standards and Technology (NIST) when President

Reagan signed into law the Omnibus Trade and

Competitiveness Act.
Library of Congress Catalog Card Number: 89-600736

National Institute of Standards and Technology Special Publication 250-27

Natl. Inst. Stand. Technol. Spec. Publ. 250-27, 308 pages (May 1989)

CODEN: NSPUE2

Certain commercial equipment, instruments, or materials are identified in this paper in order to adequately specify tlie experimental procedure. Such identification does not imply recommendation or endorsement by the National Institute of Standards and Technology, nor does it imply that

the materials or equipment identified are necessarily the best available for the purpose.

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1989

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402-9325
PREFACE

Calibrations and related measurement services of the National Institute of

Standards and Technology provide the means for makers and users of measuring

tools to achieve levels of measurement accuracy that are necessary to attain

quality, productivity, and competitiveness. These requirements include the

highest levels of accuracy that are possible on the basis of the most modern

advances in science and technology as well as the levels of accuracy that are necessary in the routine production of goods and services. More than 300

different calibrations, measurement assurance services, and special tests are available from NIST to support the activities of public and private

organizations. These services enable users to link their measurements to the

reference standards maintained by NIST and, thereby, to the measurement

systems of other countries throughout the world. NIST Special Publication

250, NIST Calibration Services Users Guide , describes the calibrations and

related services that are offered, provides essential information for placing

orders for these services, and identifies expert persons to be contacted for

technical assistance.

NIST Special Publication 250 has recently been expanded by the addition of

supplementary publications that provide detailed technical descriptions of

specific NIST calibration services and, together with the NIST Calibration

Services Users Guide , they constitute a topical series. Each technical

supplement on a particular calibration service includes:

specifications for the service

design philosophy and theory

description of the NIST measurement system

NIST operational procedures

measurement uncertainty assessment , error budget

systematic errors random errors

NIST internal quality control procedures

The new publications will present more technical detail than the information

that can be included in NIST Reports of Calibration. In general they will

also provide more detail than past publications in the scientific and

technical literature; such publications, when they exist, tend to focus upon a particular element of the topic and related elements may have been published

in different places at different times. The new series will integrate the

description of NIST calibration technologies in a form that is more readily

accessible and more useful to the technical user.

iii
The present publication, SP 250-27, NIST Measurement Services: AC-DC

Difference Calibrations, by Joseph R. Kinard, John R. Hastings, Thomas E. Lipe , and Clifton B. Childers, is one of approximately 20 documents in the new

series published or in preparation by the Center for Basic Standards. It

describes calibration technology and procedures utilized in connection with

NIST Service Identification Numbers from 53310 to 53343 listed in the NIST

Calibration Services Users Guide . Inquiries concerning the contents of these

documents may be directed to the authors or to one of the technical contact

persons identified in the Users Guide (SP 250)

.

Suggestions for improving the effectiveness and usefulness of the new series

would be very much appreciated at NIST. Likewise, suggestions concerning the

need for new calibration services, special tests, and measurement assurance

programs are always welcome.

Joe Simmons, Acting Chief

Office of Physical Measurement Services

Katharine B. Gebbie, Acting Director

Center for Basic Standards

iv
INTRODUCTION

The primary purpose of this publication is to collect and sununarize the

specialized information needed to operate the ac-dc difference laboratory and

calibration service at NIST (Gaithersburg) . This document should be used in

conjunction with the computer programs stored on discs and listed in Room A144

Metrology Building and the laboratory notebooks located in the same room.

An additional purpose, however, is to provide the users of this calibration

service and other interested people with a convenient reference source. The

service and its underlying background are documented here in considerable

detail

.

Contained in this publication are:

an annotated table of contents, a topical index, and a glossary of common

ac-dc acronyms

an overview of the service

selected published papers

instructions for the operation of the comparator systems

a schedule for the recalibration and periodic checks of the NIST thermal

converters

a sample report of calibration

The annotated table of contents and the topical index serve as a "road map" to

the different sections of this document, which consists largely of previous

writings. Each section is numbered consecutively, and the section numbers are shown on the first page of each one. The glossary lists and defines some frequently used acronyms. A good starting place in learning about ac-dc

difference calibrations at NIST is the overview presented as section 1, to

supplement the brief account given in NIST Special Publication 250, NIST

Calibration Services Users Guide .

In recent decades many papers or reports have discussed various aspects of ac- dc difference measurements, especially as they are made at NIST. Twelve of

these publications from 1952 to 1987, including one by a non-NIST author, are presented as sections 2 through 13. Collectively they document both the

theoretical and the practical basis of the NIST calibration service for ac-dc

differences. Other publications could have been added, but these 12 cover the

subject. Hence, the aids mentioned above for finding the discussions of

different topics have been prepared instead of newly written sections that

would have duplicated what already existed.

V
Sections 14, 15, and 16 are instructions for operating the comparator systems.

They were prepared for use by the NIST staff and have not been reviewed as thoroughly as the preceding sections. Nevertheless, they can be helpful to

non-NIST readers interested in operational details. They are revised from

time to time, of course, as changes are made in the equipment and procedures.

A schedule for the recalibration and periodic checks of the NIST thermal

converters follows as section 17. These quality control procedures grew out

of recommendations given in section 4. Finally, section 18 is a sample report

of calibration.

vi
Annotated Table of Contents

"AC-DC Difference Calibrations at NBS , " by J . R.

Kinard, in Proc . Meas . Sci. Conf

.

. Irvine, CA, Jan.

1986, pp. 3-8. An overview of the NBS calibration

service for thermal converters, including

descriptions of the primary, reference, and working

standards. The upper voltage and current limits and

the uncertainties of the ac-dc difference

calibrations are given as a function of frequency.

"AC -DC Comparators for Audio -Frequency Current and

Voltage Measurements of High Accuracy," by F. L. Hermach, IEEE Trans. Instrxam. Meas. . IM-25, pp. 489-

494, Dec. 1976. A survey of ac-dc transfer

instruments and basic standards.

"AC-DC Transfer Instruments for Current and Voltage

Measurements," by F. L. Hermach, IRE Trans

.

Instrum. . 1-7, pp. 235-240, Dec. 1958. Describes

different types of instruments for the measurement

of ac current and voltage and outlines the

advantages of the electrothermal type.

"An Investigation of the Uncertainties of the NBS

Thermal Voltage and Current Converters," by F. L.

Hermach, Natl. Bur. Stand. (U.S.), Rep. NBSIR 84-

2903, Apr. 1985. This report discusses: the

establishment of a new set of NBS primary ac-dc

standards; the construction of a precise comparator

for intercomparing these standards; the NBS

techniques for frequency, voltage, and current

extension; and the study and evaluation of random

and systematic errors for the NBS standards and the

calibration service.

"Multifunction Thermal Converters as the NBS Primary

AC -DC Transfer Standards for AC Current and Voltage

Measurements," by F. L. Hermach, J. R. Kinard, and

J. R. Hastings, IEEE Trans. Instrum. Meas. . IM-36,

pp. 300-306, June 1987. A conference report based

on the work reported in (4) and including later

results

.

"An Investigation of Multij unction Thermal

Converters," by F. L. Hermach and D. R. Flach, IEEE

Trans. Instrum. Meas. . IM-25, pp. 524-528, Dec.

1976. A study of a group of multifunction thermal

converters (MJTC's) which later became NBS primary

ac-dc s tandards

.

vii
page

(7) "Theoretical Analysis of the AC/DC Transfer 130

Difference of the NPL Multijunction Thermal

Converter Over the Frequency Range DC to 100 kHz,"

by F. J. Wilkins, IEEE Trans. Instrum. Meas

.

. IM-21,

pp. 334-340, Nov. 1972. Provides a theoretical

analysis of the thermoelectric and reactive errors

of MJTC's and supports the expectation of very small

ac-dc differences for the NBS primary standards.

(8) "Thermal Converters for Audio -Frequency Voltage 137

Measurements of High Accuracy," by F. L. Hermach and

E. S. Williams, IEEE Trans. Instrum. Meas. . IM-15,

pp. 260-268, Dec. 1966. Describes the construction

of the NBS Model F-1 reference set of thermal

voltage converters (TVC's). Gives the results of a study of the F-1 TVC set and special NBS

thermoelements (TE's) used as thermal current

converters (TCC's).

(9) "Thermal Voltage Converters and Comparator for Very 146

Accurate AC Voltage Measurements," by E. S.

Williams, J. Res. Natl. Bur. Stand. (U.S.) . 75C, pp.

145-154, July-Dec. 1971. Describes the construction

of the NBS Model F-7 TVC set of working standards

and the NBS manual comparator

.

(10) "Thermal Voltage Converters for Accurate Voltage 156

Measurements to 30 Megacycles Per Second," by F. L.

Hermach and E. S. Williams, Trans . AIEE

(Communication and Electronics) . 79, pt. I, pp. 200-

206, July 1960. Describes the construction and

characteristics of the NBS Model-C TVC set used from

100 kHz to 1 MHz.

(11) "Thermal Converters as AC-DC Transfer Standards for 163

Current and Voltage Measurements at Audio

Frequencies," by F. L. Hermach, J. Res. Natl. Bur.

Stand. (U.S.) . 48, pp. 121-138, Feb. 1952. An

experimental study of thermal converters as well as

a theoretical discussion of thermal transfer errors.

(12) "Thermal Current Converters for Accurate AC Current 181

Measurement," by E. S. Williams, IEEE Trans.

Instrum. Meas. . IM-25, pp. 519-523, Dec. 1976.

Describes TCC's and intercomparison techniques.

vlii
page

(13) "A Dual-Channel Automated Comparator for AC-DC 186

Difference Measurements," by E. S. Williams and J.

R. Kinard, IEEE Trans. Instrum. Meas

.

. IM-34, pp.

290-294, June 1985. Describes the NBS automated

comparator systems used for ac-dc difference

calibrations

.

191

(14) "General Instructions for the Operation of NBS

Automated AC-DC Comparator System II," by T. E. Lipe. Provides a general description and setup

instructions and tells how to: use the system

status utility and amplifier gain checking utility,

create and modify files containing calibration

procedures for the test instruments and a correction

table for the standards, calibrate various types of

thermal converters, and prepare ac-dc test reports.

(15) "Instructions for Using the Manual AC-DC Difference 262

Comparator System," by E. S. Williams and C. B.

Childers. Provides a general introduction and

describes the power supplies and controls, console,

TE comparator, and galvanometer circuits and

shields. Also discusses ac-dc difference tests,

grounding, the calculation of nE (where n is the

exponent relating the input current I to the output

EMF E = kl'^) , and low- frequency tests.

(16) "Instructions for Using the MJTC Comparison 279

Equipment," by F. L. Hermach. Discusses the ac-dc

standards, comparison equipment, n-compensation, and

the basic equation for this comparator. Describes in

detail the operation of the equipment under the

headings: connections, preliminary adjustments, n- measurement , n-compensation, dc reversal

compensation, ac setting, ac-dc determinations,

recording and computing data, and shutting down.

Also discusses the maintenance of the equipment

through inspections and periodic tests.

(17) "Schedule for Recalibration and Periodic Checks of 292

NBS Thermal Converters." Specifies regular tests of

the reference and working standards.

ix
page

(18) "Report of Calibration." A sample report. 294

NOTE: Section (4) in its original form included reprints of three

papers to make it easy to refer to them. Since these papers are also included in the present compilation as sections (6) , (8) , and

(9) , they are omitted from section (4) to avoid duplication.

X
Topical Index

Topic Sections

Specifications for the Service 1,18

An overview of the calibration service, including the frequency

dependence of the upper voltage and current limits and the uncertainties of

the calibrations, is given in (1). Section (18) is a sample calibration

report

.

Design Philosophy and Theory 2,3,4,5,6,7,8,9,10,11,12,13

A survey of ac-dc transfer instruments and basic standards is presented

in Section (2) . The advantages of electrothermal instruments are discussed in

Section (3). MJTC's are introduced in Sections (4), (5), and (6). Section

(4) is a internal report that serves as a basis for much of the calibration

service. A theoretical error analysis for MJTC's is given in Section (7).

The NBS reference and working sets of TVC's are described in Sections (8),

(9), and (10). Section (11) is noteworthy for its theoretical discussion of

thermal transfer errors. A set of TCC's is described in Section (12), and the

NBS automated comparators are discussed in Section (13).

Description of the NBS Measurement 1,4,5,6,8,9,10,13,14,15,16

Most of these sections have been cited earlier, but they are relevant to

this topic also. The primary, reference, and working standards are described

in Section (1). The most precise of the NBS comparators is discussed in

Sections (4), (5), and (6). Special NBS TE's used as TCC's are described in

Section (8), and the NBS manual comparator is discussed in Section (9).

Section (10) deals with the high-frequency TVC's. Section (13) is about the

automated comparators. Detailed descriptions of the different comparator

systems are given in Sections (14), (15), and (16).

NBS Operational Procedures 14,15,16

These references explain how to use the different comparator systems.

Measurement Uncertainty Assessment 4

A detailed analysis of the random and systematic uncertainties for the

NBS standards and the calibration service is presented in this basic

reference

.

NBS Internal Quality Control Procedures 4,17

Recommendations for maintaining the quality of the standards and the

calibration process are given in Section (4) . A schedule for recalibration

and periodic checks of the standards is presented in Section (17).

XI
Glossary of Common AC-DC Acronyms

AC: alternating current

AV: alternating voltage

DC: direct current

DV: direct voltage

TE: thermoe lement

TC: thermal converter

TCC: thermal current converter

TVC: thermal voltage converter

SJTE: single- junction thermoelement

SJTC: single- junction thermal converter

MJTC: mult ij unction thermal converter

xii
Section 1

AC-DC DIFFERENCE CALIBRATIONS AT NBS*

Joseph R. Kinard

Electricity Division

National Bureau of Standards

ABSTRACT

The NBS (Gaithersburg) calibration service for thermal voltage and current

converters relies on a group of primary multij unction thermal converters and

sets of reference and working standards for extending their ranges and

frequencies. The converter sets that constitute the NBS standards -- primary,

reference, and working -- as well as the build-up and bootstrap techniques

used in their characterization over the full ranges of voltage, current, and

frequency are described briefly. The upper voltage and current limits and the

uncertainties of the ac-dc difference calibrations are given as a function of

frequency

.

INTRODUCTION

Alternating voltage or current is commonly measured by using a thermal

converter to compare the heat produced by the alternating signal to the heat

produced by a dc signal of nearly equal magnitude. NBS (Gaithersburg)

provides a calibration service for the ac-dc differences of such thermal

voltage converters (TVC's) and thermal current converters (TCC's). (TVC

calibrations at frequencies above 1 MHz are performed at the NBS branch in

Boulder.) The ac-dc difference is conventionally defined as

5 = (Qa - Qd)/Qd

where Q^j is the average of the two polarities of dc input that produce the same output from the thermal converter as that produced by the ac input Q^.

Thermal converters are calibrated at NBS by comparing them with standards

whose ac-dc differences have been carefully determined. The NBS primary ac-dc

transfer standards are a group of multijunction thermal converters (MJTC's)

from several sources, manufactured at different times and in different ways.

Voltage, current, and frequency ranges have been extended beyond those of the

MJTC's by using reference and working sets of TVC's and TCC's.

*An earlier version of this paper was presented at the Measurement Science

Conference in January 1986. This revised version was prepared in December

1987.

1
MULTIJUNCTION THERMAL CONVERTERS

Over the past several years a series of comparisons has led to the

establishment of a group of eight MJTC's as the NBS primary standards for ac- dc difference [a]-[c]. These converters are constructed with bifilar heaters

and as many as 200 or more thermocouples to sample the heater temperature

essentially along its entire length. Thermoelectric errors, such as Thomson

and Peltier heating and cooling, as well as other errors have been shown

theoretically to be below the 1 ppm level [d] . Dielectric loss and other

defects do exist, however, but compensation techniques have been found to

greatly reduce their effect [a]. The fact that the MJTC's were obtained over

a period of years from different manufacturers who used different designs and

fabrication techniques considerably improves the confidence in this group as primary standards, since the likelihood that a systematic error will be

discovered is much greater for intercomparisons of such varied artifacts than

for repeat measurements on one standard or for comparisons of like standards.

The MJTC's have heater ratings ranging from 5 mA to 50 mA with voltage drops

from 2 V to 10 V and output EMF's of 30 mV to 120 mV. This high output EMF

and a carefully constructed comparator enable a pooled standard error (standard deviation of the mean) for MJTC intercomparisons of about 0.2 ppm to

be obtained. Based on these comparisons and the above mentioned experimental

and theoretical considerations, the average ac-dc difference of this group is

less than 0.5 ppm from 30 Hz to 10 kHz.

The extension of the voltage, current, and frequency ranges beyond those of

the MJTC's has been achieved by the use of TVC's and TCC's whose ac-dc

differences have been shown, as described below, to be nearly independent of

voltage or current level and only slightly dependent on frequency.

THERMAL VOLTAGE CONVERTER STANDARDS

The NBS reference and working TVC standards are of the coaxial single-range

type. They include special thermoelements (TE's) with Evanohm heaters and

temperature compensation [e] as well as some conventional standard pattern and

UHF type TE's. The reference and working sets of TVC's were constructed so that adjacent ranges could be compared to step up or step down on the voltage

scale. The reference TVC set (NBS F-1) consists of two 5 mA TE's and 13 plugin coaxially mounted resistors [f]. Each resistor, in conjunction with one of

the TE's, is dedicated to a particular voltage range. The extension from the

base 10 V range up to 1000 V and down to 0.5 V requires 14 adj acent- range

comparisons at each frequency. The working TVC set (NBS F-7) consists of a 2.5 mA TE, a 5 mA TE, and six plug-in coaxially mounted resistors [e] . Each

resistor may be used for two voltage ranges, since the ac-dc differences of

the resistors for both of these well constructed sets (F-1 and F-7) are nearly

independent of voltage level. For the working TVC set (F-7) only a seven- step voltage build-up and build-down comparison process is required to

characterize all the ranges from 1 V to 1000 V at each frequency.

2
The extension of the frequency range above 100 kHz was achieved by the use of

converters (NBS C-set) made with cylindrical, deposited-carbon resistors

mounted coaxially in metal cylinders with 5 mA or 10 mA UHF type TE's [g] . Two techniques were used to establish the frequency variation of ac-dc

difference up to 1 MHz. Firstly, the converters were made according to

designs whose residual reactance effects were nearly zero based on theoretical

calculations. The intercomparisons of adjacent voltage ranges, whose

resistors have very different resistances but nearly equal reactances, showed

hardly any relative ac-dc difference variations with frequency. This process

and other special tests strongly supported the theoretical calculations

indicating that the frequency coefficients were very small. Secondly, the

frequency coefficients were verified by the use of a bolometer bridge to

measure the performance of the TVC's in the 30-40 MHz region. If present, the

residual reactance error would be expected to be quite large at these high

frequencies. The bolometer bridge had sufficient accuracy to detect a reactance error equivalent to less than 100 ppm transferred down to 1 MHz. No

such error was found.

THERMAL CURRENT CONVERTER STANDARDS

The NBS reference and working standard sets of TCC's are comprised of two

types of TE's [f]. In the 5 mA to 1000 mA range the TCC's are vacuum TE's

with Evanohm heaters, some of special design. From 1 A to 20 A the current

transfer standards are air-cooled TE's made a number of years ago to NBS

specifications. Current build-up and build-down in the region up to 1 A is

accomplished for the reference standards by successively comparing two TE's of

a lower range to each other and then, after connecting them together in

parallel, to each of two TE's of a higher range. Pairs of almost identical

low-reactance resistors are connected in series with the heaters to assure

that the paralleled currents are sufficiently in phase. Above the 1 A level,

adj acent- range TCC's are compared in series at the lower rated current [h]

.

Current dependence is checked by comparing selected ranges with shunted lowrange TE's at frequencies low enough so that shunt reactance errors are negligible. All of these TE's have Evanohm heaters with small, frequencyindependent thermoelectric effects, so the measured current dependence can be

extrapolated to 50 kHz. To check for the possibility of errors from loss in

the bead insulating the thermocouple from the heater, some measurements are taken with the grounded end interchanged.

To establish the frequency dependence of the ac-dc difference for TCC's, two 5 mA TE's are compared by using them with two nearly identical 30 V series

resistors to form TVC's. By interchanging the resistors and averaging the

results, the difference in the reactances of the resistors is essentially

eliminated arid the TE's are evaluated as current converters. This method

avoids bead loss errors since both TE outputs are grounded. By using this

method to compare their frequency dependence [g] , reference TE's of different

design and construction are characterized up to 50 kHz. The frequency

independence of their relative ac-dc difference strongly indicates the absence

of reactive and other effects.

3
CALIBRATION METHODS AND UNCERTAINTIES

The calibration of customer's TVC's and TCC's is performed by comparing the

test unit to NBS standards whose ac-dc differences have been characterized as described above. In the case of voltage, the test and standard TVC's are connected in parallel to the same voltages in a sequence such as ac , dc+, dc-, ac , etc. In the case of current, the test and standard TCC's carry the same current in series. The heater and thermocouple of the test converter each

have one end connected to ground. For both voltage and current, the TE

outputs are monitored by comparator circuits that enable the difference in

response to ac and dc to be recorded. Three types of comparators are used in

NBS calibrations. In the first circuit, the output of the test TE is

monitored and held constant for applied ac and dc [e] . Simultaneously, the

difference between test and standard outputs is determined by resistively

dividing down the larger for comparison to the smaller. The second

comparator, used for the highest precision work, is a modified version of the

first [a]-[c]. Important modifications include a compensation circuit, which

relaxes the need to maintain a constant output from one TE, and an analog

integrator on the output of the null detector. The third comparator is a dual-channel type used in the NBS automated systems [i]. In this comparator,

two separate voltage nulling and monitoring circuits permit the measurement of

the variations in the output of each TE when the input is switched back and

forth from ac to dc+ or dc-. Though difficult to operate manually, this

method lends itself to automation.

Table I summarizes the frequency dependence of the upper voltage and current

limits as well as the uncertainties of the ac-dc difference calibrations at

NBS (Gaithersburg) as they exist in December 1987. These uncertainties were recently reduced, as noted at several points in the table, and further

reductions are anticipated as a result of research that is presently under

way. Figs. 1-6 show these uncertainties pictorially. Fig. 1 is for multirange TVC's and Figs. 2-4 are for coaxial single -range TVC's. The upper

voltage limit is 1000 V from 20 Hz to 100 kHz and 50 V or 100 V at lower and

higher frequencies (between 2 Hz and 1 MHz). Figs. 5 and 6 are for TCC's, for

which the upper current limit is 18 A from 100 Hz to 5 kHz and less outside

this range

.

4
REFERENCES

[a] Hermach, F. L. and Flach, D. R. , "An Investigation of Multifunction

Thermal Converters," IEEE Trans. Instrum. Meas

.

. Vol. IM-25, Dec. 1976,

pp. 524-528.

[b] Hermach, F. L. , "An Investigation of the Uncertainties of the NBS Thermal

Voltage and Current Converters," NBS Internal Report 84-2903, April 1985.

[c] Hermach, F. L. , Kinard, J. R. , and Hastings, J. R. , "Multij unction

Thermal Converters as the NBS Primary AC-DC Transfer Standards for AC

Current and Voltage Measurements," IEEE Trans. Instrum. Meas. . Vol. IM36, June 1987, pp. 300-306.

[d] Wilkins, F. J., "Theoretical Analysis of the AC/DC Transfer Difference of

the NPL Multij unction Thermal Converter Over the Frequency Range DC to

100 kHz," IEEE Trans. Instrum. Meas. . Vol. IM-21, Nov. 1972, pp. 334-340.

[e] Williams, E. S., "Thermal Voltage Converters and Comparator for Very

Accurate AC Voltage Measurements," J. Res. Natl. Bur. Stand. . Vol. 75C,

July-Dec. 1971, pp. 145-154.

[f] Hermach, F. L. and Williams, E. S., "Thermal Converters for AudioFrequency Voltage Measurements of High Accuracy," IEEE Trans. Instrum.

Meas

.

. Vol. IM-15, Dec. 1966, pp. 260-268.

[g] Hermach, F. L. and Williams, E. S., "Thermal Voltage Converters for

Accurate Voltage Measurements to 30 Megacycles Per Second," Trans . AIEE

(Communication and Electronics) . Vol. 79, Pt. I, July 1960, pp. 200-206.

[h] Hermach, F. L. , "Thermal Converters as AC-DC Transfer Standards for

Current and Voltage Measurements at Audio Frequencies," J. Res. Natl.

Bur. Stand. . Vol. 48, Feb. 1952, pp. 121-138.

[i] Williams, E. S. and Kinard, J. R. , "A Dual-Channel Automated Comparator

for AC-DC Difference Measurements," IEEE Trans. Instrum. Meas. . Vol. IM34, June 1985, pp. 290-294.

NOTE: These references are designated by letter rather than number to avoid

confusion with the section numbers of the compilation. This paper is section

(1). The following table shows the correspondence between references and

sections

.

Reference: abcdef g h i

Section: 6 4 5 7 9 8 10 11 13

5
TABLE I. AC-DC DIFFERENCE CALIBRATION SERVICE

Frequency

2-5

rlZ 5-20

Hz

20-20k

Hz

20-50

KnZ

50-100 0.

KnZ

1-0.5 C.5-1

nnZ

Voltage Limits(V) 50 100 1000* 1000 1000* 100 100

Current Limits(A) 0.05 0.05 16

UNCERTAINTY (parts per mi llion)l^b]

Multirange TVC's

> 100 V 50 70 100

< 100 V 200 100 30* 50* 70* 100* 200*

Coax Single Range

TVC's > 100 V C \J 30 50

5 V < X < 100 V 200 100 15[c]* 25* 40* 100* 200*

< 5 V 200 100 20 30 50 100* 200*

Speciall^^^

5 V ^ X < 100 V 10

TCC's

> 5 A 100 150*

50 mA < X ^ 5 A 50 70*

^ 50 mA 200 100 50 70*

Indicates reduced uncertainty or extended voltage range, compared to the

previous NBS calibration service.

^^^5 A at 20 Hz, increasing to 1 8 A from 100 Hz to 5 kHz; 16 A above 5 kHz.

20 A shunts are calibrated at less than rated current.

lower uncertainty applies at the crossover frequencies. Uncertainties

may be increased if the ac-dc differences are large or affected by selfheating or other instability.

f^c]20 ppra from 20 Hz to 100 Hz, 15 ppm at 100 Hz and above.

t<^^Normally available by prearrangement for coaxial, single range TVC's

between 100 Hz and 20 kHz. An additional cost and a longer turnaround time

at NBS are required.

6
225

200

^1175

CL ^ 150

^125

c- 100 -

rd :

QJ , 50 - C

25 r 0

IHz

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

J—I I I 1 1 III I I I II I III I I I I II III I I I 1 1 1 III

+

+

+

+

I L-t

I

I +

I I I I I III! I I I I I I I ll 10Hz 100Hz IkHz 10kHz

F requency

100kHz IMHz

Figure 1. Uncertainties for Multirange

Thermal Voltage Converters

V > 100 Volts

V < 100 Volts

E

Q.

C

-p

225

200 -

- + + + + +

175

- + + + + + 150 + + + + +

125

- + + + + +

100 + + + + +

75 r + + + + + 50 + + + +

25 + + + + 1 r+

IHz 10Hz 100Hz IkHz 10kHz

Frequency

100kHz IMHz

Figure 2. Uncertainties -For Coaxial

Thermal Voltage Converters

V > 100 Volts

7
10Hz 100Hz IkHz 10kHz

F requency

100kHz IMHz

Figure 3. Uncertainties -for Coaxial

Thermal Voltage Converters

5 V £ 100 V

Spe c i a 1 Te s

t

£

Q.

a

c

fd

225

200 + + + + +

175 + + + + + 150 + + + + +

125 + + + + + 100 —1 + + +

75 + + + + 50 + + + +

25 + + + + 1 +

0 - 1 1 i 1 1 1 III 1 1 1 1 1 1 III 1 1 II 1 Mil 1 1 1 i 1 1 III 1 1 1 Mini 1 IIIU-U4

IHz 10Hz 100Hz IkHz 10kHz

Frequency

100kHz IMHz

Figure 4. Uncertainties -For Coaxial

Thermal Voltage Converters

V < 5 Vo 1 1 s

8
225

200 >—

E

CL 150

125

c- 100

+>

j_u 50

c

75

25 -

0

IHz

I I I

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+

+H

+

I -H

+

I I I I I I I 1

1

I I I I I I I 1

1

I I I I I I I 1

1

I I I I I I I ll 10Hz 100Hz IkHz 10kHz 100kHz

Frequency

Figure 5. Uncertainties -for Thermal Current Converters

• ^ 2. ^ flmps

50 mfl < I < 5 Rmps

225

IHz 10Hz 100Hz IkHz 10kHz 100kHz

F requency

Figure 6. Uncertainties for Thermal Current Converters

I < 50 mflmps

9
IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-25, NO. 4, DECEMBER 1976 489

Section 2

AC-DC Comparators for Audio-Frequency Current and

Voltage Measurements of High Accuracy

FRANCIS L. HERMACH, FELLOW, IEEE

Abstract—This paper surveys recent developments in high- accuracy ac-dc comparators, the basic standards for ac current and voltage measurements at audio and higher frequencies. These

instruments compare the unknown ac quantity with a dc reference. The ac-dc transfer characteristics of thermal converters, the most

widely used comparators, can be evaluated to about 10 ppm at audio

frequencies in national metrology laboratories. With recent de- velopments even better accuracies, and greater convenience with

automatic comparisons, are feasible.

Basic Principles

IT IS NOW FAIRLY well known that all measure- ments of ac current and voltage depend fundamentally

upon precise ac-dc comparators (ac-dc transfer standards), which either simultaneously or sequentially compare the unknown ac quantity with a known equal or nearly equal dc counterpart. Such comparators are designed and constructed for high precision and small stable

ac-dc differences,^ but need not have the long term stability of reading required in other instruments. They are used with basic dc standards to evaluate, for example,

calibrated ac voltage or current sources. These in turn are used to evaluate instruments such as digital voltmeters.

Manuscript received June 29, 1976. The author is with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

^ The ac-dc difference is defined as 6 = (Qar ~ Qdc)/Qdc. where Qac and

Qdt are the ac and dc quantities (current or voltage) required to produce

the same response (output) of the comparator.

or ac-dc converters for such instruments. Since accuracies

of commercially available sources and instruments now

approach 0.01 percent (100 ppm), very high accuracies are required in the ac-dc transfer. The determination of the ac-dc differences of such

comparators has been, and continues to be, an important

task of national metrology laboratories. The ampere or the

volt is established with direct current or voltage in these

laboratories to a few ppm by electrodynamic or electrostatic instruments, which in effect weigh the electrical unit

by balancing an electrical force against the force of gravity.

The pertinent constants of the instruments must be determined to the full accuracy of the measurement. While

in principle this could be done with alternating current or

voltage, the difficulties would be greater. The most re- producible or stable references are based on the Josephson

effect and the cadmium standard cell, which provide dc

standards. For these reasons ac measurements are referred

to these dc standards by ac-dc transfer standards whose

constants must simply be equal at the two frequencies.

Either the rms, crest, or average value of an ac wave may

be compared with a dc value. The rms value is most often

desired or required, because it governs the interchange of

electrical with other forms of energy. However one of the

others may sometimes be more suitable, as in iron-loss oc

dielectric strength testing.

One of these values cannot be deduced readily from

another. This is not academic. As accuracies increase, the

10
490 IEEE TRANSACTIONS ON INS TKUMENTA riON AND MKASUHEMKNT. DECKMBKR 1976

common error of measuring a voltage or current of un- known waveform with an average- or peak-responding

instrument scaled in terms of the rms value of a sine wave

can become very serious. Because of their simplicity such

instruments are now widely used. To avoid compounding

the error they should be calibrated with corresponding

peak-to-dc or average-to-dc comparators, or with an ac source having a crest factor or form factor (as needed)

equal to that of a sine wave to the required accuracy. Highly accurate peak and average ac-dc comparators have

been designed, but only over limited ranges to date. RMS-DC Comparators

Classical methods of ac-dc comparison are based on

electrodynamic, electrostatic, or electrothermic instruments, and have been described in earlier survey papers

[l]-[4]. Each of these comparators responds inherently to the rms value of the voltage or current because of its operating principle and its mechanical or thermal inertia. For example, the equation governing a conductor heated

by a current i with all of the conductor at the same temperature is, at each instant, i'^R = kd -\- Mdd/dt, where R,

k, and M are parameters which in general depend upon the

temperature rise 0. If the thermal inertia is high enough so that 6 is essentially constant during the cycle then, by

termwise integration over one cycle, the steady-state

temperature rise is simply da - I^R/k, where / is the rms

current. The temperature rise is thus a true measure of the rms

current / even if R and k are temperature dependent. This

inherent rms response holds under much more general

conditions as long as the cyclic variations of 6 are negligible.

Note that as an ac-dc comparator, da does not have to be an exact quadratic function of /. However, at low

frequencies the average value of the cyclically varying d is then not correctly a measure of /. Similar relationships hold for electrodynamic and

electrostatic instruments, but have not yet been synthesized with high accuracy by electronic means.

Thermal Converters

The temperature rise of a thin straight conductor

carrying a given current is nearly independent of frequency

over a wide range because of the low reactance of the wire,

and can easily be sensed with high precision by one or more

thermocouples. The ac-dc differences of such thermal

converters (TC's) are quite small and have been studied

extensively in national laboratories over the past few

decades. TC's with appropriate electrical "readouts" have

almost entirely replaced the other forms of ac-dc comparators, which are at present more limited in accuracy by

frequency-dependent effects or in ranges of voltage and

current. A conventional simple TC (in ranges from about 2 to 500

mA), consists of a small evacuated glass bulb containing

a heater wire V2 to 1 cm long, with the hot junction of a

single thermocouple thermally connected to its midpoint

by a tiny insulating bead. Thomson and Peltier heating

and cooling of the heater can affect the temperature on

direct current.^ Even when the average of the two directions is taken a small error remains. Because of thermal

inertia they do not affect it on alternating current (except

at low frequencies), leading to a frequency-independent

ac-dc difference [5], [6]. Fortunately this can be reduced

to a few ppm or less by proper design and selection of

materials. However, exact calculation is questionable, even though appropriate formulas have been developed, be- cause of possible effects of heat treatment on the ther- moelectric constants during fabrication.

Many commercial TC's have lead-in wires of magnetic

alloys to match the temperature coefficient of the glass

bulb. Skin effect in these leads can cause additional ac

heating. Even more serious ac-dc differences can occur in voltage measurements because of the increased ac resis- tance. They are minimized with nonmagnetic leads, which,

however, must have low thermal EMF's versus the heater

alloy and yet provide a good seal. Dielectric loss in the insulating bead generally varies

with frequency and temperature, and can thus cause significant ac-dc differences. The differences are dependent

on current, frequency, and the voltage between the heater

and thermocouple.

Thermoelectric errors are negligible in multijunction

thermal converters (MJTC's) in which many series-connected thermocouples are spread along the heater. Very

ingenious designs have been developed, some with as many

as 200 junctions and with bifilar heaters which further

reduce these errors to theoretical values far below 1 ppm

[7]. They have other advantages as well, such as higher

rated output (up to 100 mV instead of 10). Investigations

have shown, however, that some constructions are subject

to dielectric loss and other errors [36]. Series resistors are used with TC's to form thermal

voltage converters (TVC's). These may be wire-wound or film-type resistors, switch-selected for convenience in obtaining multiple ranges, typically from 1 to 1000 V.

Cylindrical metal-fiLm resistors mounted coaxially in metal

cylinders make it possible to compute the effects of re- sidual reactances to 10 MHz or more, at least at low volt- ages. Intercomparisons of adjacent ranges of such sets of

TVC's provide valuable confirmation and also assurance

against other sources of ac-dc differences (which are not

likely to be the same for all ranges). Such tests can be made

to a few ppm at audio frequencies [8]. With two TC's of different current ratings (for example,

2.5 and 5 mA), a single series resistor can serve for two

voltage ranges. With only 6 resistors in a typical set, ranges

from 1 to 1000 V can be obtained. If they are properly designed and constructed, only 7 intercomparisons are necessary to determine their relative ac-dc differences [9]. Similarly, shunt resistors can be used with two TC's of

different voltage ratings to make convenient thermal

^ They also cause dc reversal differences when the bead is not at the exact thermal midpoint.

11
HERMACH: AC-DC COMPARATORS OF HIGH ACCURACY 491

. y .

Fig. 1. Simultaneous ac-dc comparator.

current converters (TCC's), with ranges up to 20 A or more. Transformers can also be used to provide voltage ranges. Only the heater of the TC needs to be switched to the dc

reference. Excellent accuracy is possible at midaudio

frequencies, but the design of suitable transformers for wide voltage and frequency ranges is difficult. Simultaneous ac-dc comparators have been developed,

to reduce drifts from ambient and self-heating temperature changes, which can sometimes be serious. These

usually have dual circuits, as shown for one such arrangement in Fig. 1 [10]. One TVC is supplied by the ac voltage

and the other by a known dc voltage with the TVC outputs

connected in opposition to a null detector. Unfortunately

the input-output characteristics of TC's are not equal, (do

not "track") even when equalized at one level. Thus for highest accuracy, a preliminary adjustment to null the

detector with both converters connected to the same cur- rent or voltage is generally required. This is accomplished

with the adjustable resistor shown in Fig. 1, which forms

a voltage divider with the internal resistance of the thermocouple (not indicated).

Alternatively, the null-balance adjustment can be made

at a fixed dc current level, which is supplied to both heaters

by a Zener-diode reference. The heater of one TC is then

switched to a decade range resistor (or an amplifier with

gain-control resistors) for the ac measurement. With

proper choice of components the resistor can be direct

reading in volts when it is adjusted to obtain the same

output balance [11], [12]. Similar arrangements can be used with indirectly heated

thermistors, each consisting of a small thermistor bead in thermal contact with the heater wire. Appropriate bridge

circuits have been developed for the output balances. High

sensitivity is obtainable because of the high negative

temperature coefficients of the thermistors [13], [14]. Such simultaneous comparators, with TC's or thermistors, can provide known fixed ac reference currents or voltages for ac potentiometers (and are sometimes called "ac standard cells"). In a true potentiometer an unknown

voltage (or a definite fraction of it) is balanced against an

adjustable known voltage through an appropriate null

Output EMF's can differ by several percent with single junction TC's and 0.1 percent with MJTC's.

detector. For ac measurements the two voltages must be

of exactly the same frequency, magnitude and phase. Thus

the known voltage must be phase-locked to the unknown

and must be adjustable with high accuracy without af- fecting the stability of the reference. In addition, with the

usual tuned detector, only the fundamental component of

the unknown is measured. For these reasons ac potentiometers have been used in the USA chiefly for special

measurements, when both phase and magnitude information are required.

The heater of a TC is easily burned out by overloads.

This serious disadvantage can now be overcome, however,

by rapidly acting relays or electronic circuits, which are

available with some comparators [15]. This is very desirable, provided that normal changes in the components do

not affect the ac-dc difference.

The continuing development of ac-dc comparators

makes standardization of accuracy classes, voltage and

frequency ranges, etc, impracticable. However, stan- dardization and agreement on terminology (including

accuracy), general requirements, and test methods is very

desirable, and has been accomplished in American National Standard ClOO.4 [16].

Use of Thermal Converters

For the measurement of an ac voltage, or the calibration

of an ac voltmeter, the TVC is connected as shown in Fig.

2. With Vac applied, the output EMF is balanced against

a stable dc source with a null detector (not shown). The

TVC is switched to a known dc voltage, which is adjusted

to give the same output EMF. The dc voltage is reversed

and readjusted. The detector null should be checked with

the ac voltage applied to guard against drifts. Then Vac =

Vdc(l + 1^), where h is the fractional ac-dc difference of the

TVC and Vjc is the average for the two directions.

Alternatively, if the detector is linear, its scale can be

calibrated by adjusting the dc voltage by a small known

amount AV and noting the detector change AZ). A fixed

dc voltage (nearly equal to the ac voltage) can be set as a convenient reference, and the detector reading can be re- corded at each step in a sequence such as Vjc-I-, Vac, V^c".

Then Vac = Vdc[l + 5 + s(Z) ac - Ddc)], where Dac and Ddc

are the detector readings with Vac applied and the average

of the readings with the two directions of Vjc applied, re- spectively, and s = A V/VAD. Errors from drifts are minimized if the steps are taken at equal time intervals.

The switches and output circuit should be shielded to avoid interaction. Coaxial and shielded leads should be

used as shown, to minimize induced voltages, particularly

at the test frequency. The voltage drops in the leads between the ac-dc switch and Vac and Vdc should be equal,

and errors from ground currents must be evaluated.'* Similar techniques are used for ac current measurements, but careful analysis of ground-current errors is

^ If a grounded instrument is used to measure Vdc it should be con- nected between the two switches.

12
492 IEEE TRANSACTIONS (JN INSTRUMENTATION AND MEASIIHEMRNT, IJKCEMBKR 1976

r

Fig. 2. Sequential ac-dc transfer.

Fig. .3. Basic TC feedback circuit.

needed, particularly in the test of another instrument

which must itself be grounded.

A typical commercially available comparator has a

built-in ac-dc transfer switch, and an adjustable stable dc

source in series with a detector to balance the output EMF.

Most of them do not have dc reversal switches, however,

generally relying on TC's with low dc-reversal differences.

This should be checked periodically at two or more heater

currents, for the error in the voltage measurement is ^ this difference, and can be much larger than the ac-dc differ- ence. Standard C 100.4 describes some simple yet practical

ways to detect and guard against ground-current errors, radio-frequency interference and other errors in the in- tercomparison of two current or voltage comparators.

These techniques can also be applied when ac-dc comparators are used for ac measurements or for the calibration

of ac instruments. Comparators with 3- or 4-terminal in- puts (with properly designed shields and driven guards)

greatly reduce circuit-connection errors, but are not widely

used.

If the ac-dc difference of the comparator is known and

applied as a correction along with corrections for the dc

standards, and if proper techniques are used to made valid

measurements, it is not at all necessary that the "accuracy

of the standard shall be ten times that of the test instrument" as is so often stated. The uncertainty in the ac measurement is simply the sum of the residual errors in the

standards and the random error in the test (caused by both

the standard and test instrument). In this way maximum

use is made of the inherent accuracy of the ac-dc comparator and the dc standards. The application of such cor- rections to remove determined errors avoids this 10/1 noose, which often strangles echelons of laboratories with

unnecessary accuracy requirements.

Automatic Comparators and Converters

All of these manual procedures are time consuming and

tedious. Automatically-switched comparators and ac-dc

converters (which generate a dc voltage accurately proportional to the input) have been developed to simplify the

measurements. In some automatic comparators, essentially

the same procedure is followed as in manually operated

instruments (with TC's of low dc reversal difference and

drift to simplify the ac-dc sequence). In others, the

switching is done quite rapidly (at about 1 Hz, for example)

to overcome drifts. The ac component at the switching

frequency in the detector is a measure of the difference

between the ac and dc voltages, either directly or through

a feedback circuit [17]. Gain-stabilized amplifiers provide

wide voltage ranges conveniently, with minimum loading.

For rms measurements many ac-dc converters are based on the feedback circuit shown in simplified form in Fig. 3

[18]. When /ac is applied, /dc increases until the outputs

of the two matched TC's are nearly equal. If the trans- conductance of the amplifier is G and E = kP for each TC,

/dc = Gfe (/ac^ - /dc')- Then if a = I/Gkid, « 1, /dc * Vl — a «^ /ac- Interestingly, this is an rms instrument

with a linear scale (except near zero).

To avoid tracking errors, decade input resistors, or amplifiers controlled by decade feedback resistors, can

provide appropriate voltage ranges at a fixed current. Alternatively a dual-heater MJTC can be used with fixed

resistors for tracking to about 0.05 percent over a range of

currents. Other types of converters have also been developed and

used in this general feedback circuit. One uses dual monolithic thermoelements, with two resistors and two

temperature-sensing transistors on a single chip [19]. In another, two indirectly heated thermistors, each with dual

heaters, are used with dual feedback so that each thermistor is always at the same temperature, to avoid tracking

errors [20]. These feedback circuits greatly reduce drifts and the

effective response time of the TC's (which is about 1 s). However the ac-dc differences of the input resistors and

amplifiers in these instruments must be small and stable. In addition, the design of the feedback amplifier is complicated by stability problems and by the double-frequency

ripple in the TC output at low frequencies.

A number of electronic techniques for rms to dc converters have been developed, based on approximating a square-law function with straight-line segments or with an appropriate transistor characteristic. Feedback circuits have made these more practical [21]. They have shorter

response times and wider ranges than thermal converters.

Other AC-DC Comparators

Electrodynamic and electrostatic instruments have been

used as ac-dc comparators. The frequency range of an

electrodynamic instrument, however, is limited by the

13
HERMACH: AC-DC COMPARATORS OF HIGH ACCURACY

reactances of the coils. The range can be much greater for an electrostatic instrument, but the torque/weight ratio

is low at moderate voltages, and surface films can cause ac-dc differences that are difficult to evaluate. With both

types the double-frequency periodic variations in the ac torque can cause vibrations in parts of the moving system,

with resultant errors. It is difficult, also, to obtain a reso- lution of a few ppm even with light-beam pointers. For all of these reasons these instruments find their place mainly

in national metrology laboratories, where these problems

can be studied at length and overcome [22]-[24].

Putting a rectifier in the feedback loop of a suitable

operational amplifier greatly reduces the effects of the

rectifier's imperfections. Refinements of this arrangement

have made average-responding ac-dc comparators and

converters of 20 ppm accuracy possible [25]. Instruments

with electronic or electromagnetic relays which compare

the crest (peak) value with a dc voltage have also been re- fined to better than 50 ppm, at least at low audio

frequencies [26], [27]. Rapid electronic sampling and

computing techniques, which measure the instantaneous

value of the wave at selected intervals, now make it possible to compute the rms, crest and average values, and thus to determine crest- and form- factors with high accuracy as

well [28], [29].

Evaluation of AC-DC Comparators

The ac-dc differences of well-designed and well-constructed comparators are small and quite stable. In many

cases they need not be redetermined, once known, if all adjacent ranges (such as 10 and 20 V) can be compared to detect changes. The evaluation of ac-dc differences in an

absolute sense is difficult and time consuming. It has

generally been attempted only at national metrology lab- oratories, which can carefully establish and study ac-dc

transfer standards, develop the necessary precise inter- comparison apparatus, and then use this equipment to

calibrate the comparators of many other laboratories. In

this way the standards receive wide use in establishing

compatability and accuracy in these measurements. This

is most economical, too, because the results of the research are then available to all users. Such services are important

functions of national laboratories.

These "absolute" ac-dc standards must be investigated

both theoretically and experimentally. This usually in- volves 1) theoretical analyses of sources of ac-dc errors, 2) design and construction of standards that minimize these

errors, 3) comparisons and other tests of these standards

to evaluate the errors, and 4) comparison with another type

of transfer standard which has also been carefully studied.The last step is very desirable to guard against errors from unknown sources, but it implies the development of

different types of standards of nearly equal precision and

ac-dc accuracy. At the present time other types cannot

match thermal converters in ranges or accuracy. Thus

comparisons of different kinds, constructions and ranges

493

of TC's, which are free of known sources of error, are vital. If these converters agree in ac-dc difference it is unlikely

that they have a common source of error, because of the

differences in their design and construction.

Remarkably stable ac and dc power supplies are now

commercieilly available, with rms fluctuations of only a few

ppm in a minute or so. Extraordinary stabilities (0.5

ppm/h) have been achieved in recent special laboratory

designs [30], [31]. These, together with techniques which

largely cancel the effects of drifts and of small supply

variations, have made possible high precision in the in- tercomparison of TC's and TVC's.

Careful evaluations of TVC's from 1 to 1000 V and TCCs

from 1 mA to 1 A or more have been made in national

laboratories to about 10 ppm at audio frequencies, with

reduced accuracies at higher frequencies [8], [32], [33], [34]. Comparisons with electrostatic and crest-responding in- struments in these laboratories have provided valuable

confirmation of the absence of large errors. At the Electrotechnical Laboratory in Japan an electrostatic volt- meter is planned, to provide the basic ac-dc transfer above

1 kV [35]. The development of MJTC's now makes even better

accuracy possible. At NBS MJTC's of different manufacture and range have been compared, with a standard

deviation of less then 0.2 ppm. A number of sources of error which cause MJTCs to depart from the ideal have been

uncovered, and either overcome in new designs or compensated. The ac-dc difference of a group of 8 TVC's (6 to 9.V) is believed to be less than 0.3 ppm at 160 Hz, the fre- quency of an important absolute volt determination, and

0.5 ppm from 30 to 10,000 Hz [36]. An intercomparison among several national laboratories

of 4 sets of TC's and TVC's is now almost finished. It should provide a better basis for international agreement

on the ac-dc transfer of current and voltage from 40 Hz to 50 kHz.

Conclusions

At the present time research and development have

provided ac-dc comparators which have the accuracy

needed (about 10 ppm) to support commercial developments in ac current and voltage measurements at audio

frequencies. With electronic aids, more rapid and convenient automated comparators are now feasible and are beginning to appear. There is no slackening in the importance or development of this field of measurements.

A cataloging of all these developments would be neither

feasible nor digestible in a survey paper such as this. The

basic principles of a number of them have been highlighted, instead, to illustrate the present status and recent trends in high-accuracy ac measurements.

References

The following references in the open (published) liter- ature were selected because of their general nature or ex- tensive bibliographies, or because they illustrate recent

14
494 IEEE TRANSACTIONS ON INSTRUMENTATION ANO MEASUREMENT, DECEMBER 1976

developments. A list of references published between 1945

and 1975 on high accuracy audio-frequency ac-dc current and voltage comparators is available on request from the

Electrical Instruments Section, National Bureau of

Standards, Washington, DC 20234.

M. Iwamura and T. Yamazaki, "General review of maintenance of alternating current standards," Circular No. 147, ETL, Dec. 1957.

(In Japanese with English abstract, 81 references.)

F. L. Hermach, "Ac-dc transfer instruments for current and voltage measurements," IRE Trans. Instrum., vol. 1-7, pp. 235-240, Dec.

1958.

T. B. Rozhdestvenskaya, "Electrical comparators for measurements

of current, voltage and power," JPRS 29373, 187 pp., 1965, National Technical Information Service, Springfield, VA 22161. (English

translation of a Russian Monograph with 112 references.) H. Helke, "Ac-dc converters and comparators for alternating current," Arch. Tech. Messen, no. J942-2, pp. 227-232, Oct. 1967; no. J942-3, pp. 255-260, Nov. 1967; no. J942-4, pp. 11-13, Jan. 1968. (In German, 137 references.)

F. L. Hermach, "Thermal converters as ac-dc transfer standards

for current and voltage measurements at audio frequencies," J. Res. Nat. Bur. Stand., vol. 48, pp. 121-138, Feb. 1952.

F. C. Widdis, "Theory of Peltier- and Thomson-effect errors in thermal ac-dc transfer devices," Proc. Inst. Elec. Eng., vol. 109, pt. C (Monograph 497 M) pp. 328-334, 1982.

F. J. Wilkins, "Theoretical analysis of the ac/dc transfer difference of the NFL multijunction thermal converter over the frequency

range DC to 100 kHz," IEEE Trans. Instrum. Meas., vol. IM-21,

pp. 334-340, 1972.

F. L. Hermach and E. S. Williams, "Thermal converters for audio

frequency voltage measurements of high accuracy," IEEE Trans. Instrum. Meas.. vol.TM-15, pp. 260-268, 1966.

E. S. Williams, "Thermal voltage converters and comparators for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand.,

vol. 75C, pp. 145-154, 1971.

F. L. Hermach, J. E. Griffin, and E.S.Williams, "A system for ac- curate direct and alternating voltage measurements," IEEE Trans. Instrum. Meas., vol. IM-14, pp. 215-224, 1965. W. Rump, "On the exact absolute measurement of a-c voltages and

a potentiometer for calibrating a-c instruments," Elektrotecknik,

vol. 5, pp. 64-67, 1951 (in German).

J. E. Griffin and F. L. Hermach, "A differential thermocouple

voltmeter," AIEE Trans., vol. 81-1, pp. 339-345, 1962.

F. C. Widdis, "The indirectly heated thermistor as a precise ac-dc

transfer device," Proc. Inst. Elec. Eng., vol. 103B, pp. 693-703,

1956. H. J. Schrader, "A-c current measurements with indirectly heated NTC resistors," ETZ, vol. 73A, p. 547, 1952 (in German). "IR sensor protects transfer standard," Electronics, vol. 48, pp.

[16

(17

(18

[19

[20

[21

[22

[23

[24

[25

[26

(27

[28;

[29

[30

[31

(32

[33

[34

[35

[36

36-37, Oct. 16, 1975. "AC-DC transfer instruments and converters," American National Standard, ClOO.4, 1973. ANSI, 1430 Broadway, New York, NY

10018.

L. G. Cox and N. L. Kusters, "An automatic rms/dc comparatfjr,"

IEEE Trans. Instrum. Meas., vol. IM-23, pp. 322-325, 1974. R. Gilbert and J. Miller, "Thermal converter," U.S. Patent 2 857

569, 1958. W. E. Ott, "A new technique of thermal rms measurement," IEEE

J Solid State Circuits, vol. SC-9, pp. 374-380, 1974.

P. Richman, "A new wideband true rms-dc converter," IEEE Trarui. Instrum. Meas., vol. IM-16, pp. 129-134, 1967.

P. Richman and W. Walker, "A new fa.st-computing rms/dc conversion," IEEE Trans. Instrum. Meas., vol. IM-20, pp. 313-319

1971. W. Smith and W. Clothier, "Determination of the dc-ac transfer

error of an electrostatic voltmeter," J. Inst. Elec. Eng., vol. 101, pp

465-469, 1954. T. Yamazaki, S. Iwamoto and T. Ando, "Estimation of the ac/dc

differences of electrostatic voltage comparator," Bull ETL, vol. 32, pp. 1085-1099, 1968. G. Schuster, "A high resolution electrodynamic ac-dc power transfer instrument," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 330-333,

1974.

L. A. Marzetta and D. R. Flach, "Design features of a precision

AC-DC converter," J. Res. Nat. Bur. Stand., vol. 73C, pp. 47-55,

1969.

P. Richman, "A new peak-to-dc comparator for audio frequencies,"

IRE Trans. Instrum., vol. I ll, pp. 115-122, 1962. D. R. Flach and Louis A. Marzetta, "Calibration of peak ac to dc

comparators," Proc. ISA, paper no. 14.2-3-65, 1965.

F. Deist and R. Kitai, "Digital transfer voltmeters," Proc. Inst. Elec. Eng., vol. 110, pp. 1887-1904, 1963. R. S. Turgel, "Digital wattmeter using a sampling method," IEEE

Trans. Instrum. Meas., vol. IM-23, pp. 337-341, 1974. H. Schoenwetter, "An ultra stable power supply for an absolute volt determination," A/etro/ogia, vol. 10, pp. 11-15, 1974. G. Schuster, "A precision stabilized sine wave source for AC power

measurements," IEEE Trans. Instrum. Meas., vol. IM-22, pp. 391-394, 1973.

S. Iwamoto and H. Hirayama, "Ac/dc thermal converters of the ETL," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 326-329,

1974.

C. H. Dix, "Electrical standards of measurement, Pt 1, dc and low frequency standards," Proc. Inst. Elec. Eng., vol. 122, no. lOR, pp. 1018-1036, 1975. 0. P. Galakhovaand T. B. Rozhdestvenskaya, "State of the stan- dardization of references and methods of dc and ac voltage measurements," Meas. Tech., vol. 18, pp. 1339-1341, Sept. 1975. K. Shida, "Ac-dc high voltage comparator with balance," Bull. ETL,

vol. 39, pp. 415-420, 1975 (in Japanese).

F. L. Hermach and D. R. Flach, "An investigation of multijunction thermal converters," to be published.

15
Section 3

AC-DC Transfer Instruments for Current and

Voltage Measurements*

FRANCIS L. HERMACHf

Introduction

THEbasic electrical units are maintained by groups

of standard cells and resistors. With these standards and a dc potentiometer, measurements of direct voltage and current are readily made with an ac- curacy of 0.01 per cent or better. Corresponding meas- urements of alternating voltage and current at power

and audio frequencies depend, at the present time, upon

ac-dc transfer instruments which have nearly equal ac and dc response. Such instruments may be calibrated on direct current and then used on alternating current, or alternatively, may be used directly to make ac-dc

difference measurements, as in the certification of other

instruments.

In addition to the instantaneous values as determined by an oscilloscope or point-by-point methods,

there are at least seven other quantities that may be

determined for a periodic ac wave in which all values

separated by the periodic time, T, are equal. These are:

1) the rms or effective value, /, defined by

1 c'' T Jo

where i is the instantaneous value; 2) and 3) the positive

and negative average values, i.e., the average of all the

positive or all of the negative values during one cycle,

defined by

/»+ = I (I f

I + i)dt and la- = I ( | m ~ ^)^' 2T J 0 2T J 0 where \i\ is the magnitude of i; 4) the rectified full- wave average, defined by

5) and 6) the positive and negative crest or peak values;

and 7) the crest-to-crest or peak-to-peak value.

Instruments for measuring each of these values of an

ac voltage or current wave are commercially available

(with widely varying accuracies). Ordinarily, however,

only one of these values is really required in a measurement. Unfortunately, it is difficult to deduce accurately one value from the measurement of another, except for a few waveforms (which, of course, must be known). In most cases the rms value is the one really desired, since

the rate of transformation of electrical to other forms of

* Manuscript received by the PGI, August 14, 1958.

t National Bureau of Standards, Washington, D. C.

energy (which is governed by the heating produced by

a current and by the electromagnetic force produced by

currents acting on each other) is a function of the square

of this value.

The three general types of ac-dc transfer instruments

which are at present suitable for rms measurements of better than 0.1 per cent accuracy are dependent upon

these same laws of the interchange of energy. They are:

1) electrodynamic instruments, which depend upon the

force between current-carrying conductors; 2) electrostatic instruments, which depend upon the force between charged conductors; and 3) electrothermic instruments, which depend upon some effect produced by the

heating of a current-carrying conductor. The first and

third respond essentially to current and the second to

voltage, but series or shunt resistors make all three types

suitable for both measurements, while other circuit ar- rangements make power measurements almost equally

feasible. Response Equations

The instantaneous torque of an electrodynamic in- strument in which two sets of conductors, one fixed and

one rotatable about an axis, carry the same current,

i, is^ dM

T =

dd

where M is the mutual inductance and d some welldefined angle between the two sets of conductors.

In its simplest form such as instrument has an opposing torque, Ud, so that, with direct current applied, the

rest position, d/, is defined by

dMl

Udf = BiP where Bi =

dd J

The differential equation governing the angular deflec- tion is^ Pd" + Ad' + Ud BP

where P and A are the inertial and damping constants,

respectively. If the mechanical inertia is sufficiently

great that with alternating current applied the periodic

fluctuations of 6 are negligible compared with the aver- age value, termwise integration over an integral number

of cycles gives

* F. K. Harris, "Electrical Measurements," John Wiley and Sons,

Inc., New York, N. Y., pp. 47 and 413; 1952.

16
236 IRE TRANSACTIONS ON INSTRUMENTATION December

Udf

I m = BiP.

T J 0 Thus if U and B each have the same vahie on direct as on alternating current, the instrument has the same

response, df. The instantaneous torque of the usual two-element

electrostatic instrument with an applied voltage, v, is dC ~

2 dd

where C is the capacitance between the elements. Similar considerations lead again to identical expressions

for the rest position with direct or alternating voltage

applied, viz.,

Udf = BzV- where B2

1 dC

2 With reasonable restrictions, the temperature rise, 6, of a thin homogeneous conductor carrying a current i in the x direction is governed by the differential equation

He i'R

K K

1 dd D dt

where D is the thermal diffusivity and R, K, and H are the electrical resistance, thermal conductance, and rate

of surface heat loss per degree rise of a unit length of the material. Again, if the "thermal inertia" is sufficiently great that periodic fluctuations in temperature

are negligible, the average temperatures are equal with

equal direct and rms alternating currents. In each instrument the equality holds even if the "constants" such as U, B, R, D, H, and K in the equations are dependent upon the response, 6. This important advantage stems from the property that each in- strument combines in a single measuring element a function proportional to the square of the instantaneous

current or voltage, a restraining function, and an inertia! function or "flywheel effect" to enable it to integrate

so that the time-average value of the response is proportional to the square of the rms current or voltage. For

this reason high accuracy may be more easily attained

with these instruments than with other squaring devices or circuits, which must synthesize exact and equal

square-law responses in two quadrants (or in one quadrant with an accurate rectifier) and provide a separate

integrator. However, each of these three types of instruments has a lower limit of frequency below which the

average value of the response is incorrect if the constants are dependent upon d. Transfer Instruments

It is easier in an actual instrument to insure that the constants in the equations are indeed equal on direct

and alternating current than it is to measure them sep- arately and compute the response of the instrument.

Thus with few exceptions- absolute instruments, such

as the electrodynamic current balance,' are used solely

for dc measurements, and most ac instruments of high

accuracy are calibrated in terms of dc standards. In the

usual instruments, this calibration is preserved on a

scale, and the instrument is tested or standardized periodically to guard against changes in the factors which

affect the response. However, if the instrument is cali- brated at the time of each use, long-time stability, ex- tremely low temperature coefficients, and other nor- mally desirable characteristics become of secondary importance, and the instrument may be designed and constructed to have high resolution and the best possible

frequency characteristic. Indeed, in a standardizing

laboratory in which the standard is used primarily to

test other instruments, a dc calibration may not be nec- essary because most instruments which are submitted

to such laboratories can also be used on direct as well as alternating current. The scale calibration of such instruments may be verified on direct current with a potentiometer (taking the mean of the values required to obtain the same deflection for the two directions of applied current or voltage). The ac-dc difference or frequency

influence may then be determined by connecting the

instrument under test and the transfer instrument so that they respond to the same quantity (current, volt- age, or power) and successively switching both to alter- nating-, direct-, reversed-direct, and alternating current. In each case the measured quantity is adjusted to obtain the same deflection of the test instrument and the

differences in deflection of the standard instrument are observed. From these differences and the scale factor of the standard instrument (per cent change per division)

the ac-dc difference of the test instrument may be determined.

This procedure has been used for over 40 years in the

certification of instruments at the National Bureau of Standards, because it separates several sources of errors and can be carried out with great accuracy. Since the

ac-dc differences of a well-designed instrument are small within its working frequency range and are rela- tively permanent, they need not ordinarily be redetermined, so that the periodic tests of the instrument need

be made only with direct current. For either a direct ac measurement or an ac-dc dif- ference determination, however, the ac-dc difference of the standard instrument must be known to the full ac- curacy of the measurement. The determination of this difference has been recognized as a major problem in

' H. B. Brooks, F. M. Defandorf, and F. B. Silsbee, "An absolute

electrometer for the measurement of high alternating voltages,''

/. Res. NBS, vol. 20, pp. 253-316; 1938.

' R. L. DriscoU and R. D. Cutkosky, "Measurement of current with the NBS current balance," /. Res. NBS, vol. 60, pp. 297-305;

1958.

17
1958 Hermach: AC-DC Transfer Instruments for Current and Voltage Measurements 237

most national standardizing laboratories. In general,

there are three rather distinct steps in a complete determination. These are: 1) choice of two dififerent types

of instruments that are inherently suitable as ac-dc

transfer instruments, and a theoretical study of all known effects that can cause ac-dc differences in each

type; 2) construction of instruments of each type in such a way that each of these effects by computations

and tests can be evaluated over the required ranges;

and 3) comparison of the actual t*'ansfer performance

of the two instruments to guard against unknown

sources of error. The tests of step 2) will often take three

forms. These are: a) tests of components of the instrument for the effects of known factors (such as magnetic

susceptibility, for example; b) intercomparisons of instruments of the same type but of different ranges, in which some of the effects (such as residual reactance)

may be expected to be very different; and c) repeated

intercomparisons, in which each known source of error

in one instrument is accentuated by definite amounts

(such as by the deliberate introduction of known react- ance). Normally the comparisons of step 3) can be made

over only part of the useful ranges of frequency, voltage,

etc., and the performance must also be judged by the re- sults of steps 1) and 2). Normally also one of the two in- struments will have better-known characteristics,

wider or more suitable ranges, or better-behaved ac-dc

differences than the other, and thereafter is used as the

principal transfer instrument for that laboratory, but

both are important.

The accuracy of ac measurements is often limited by

the stability of the source rather than the instrument.

For the calibration of instruments with a transfer instrument, stabilized sources free from short-time fluctuations are required, but long-time stability and good load

regulation are of less importance.'''* Repeated sets of readings make it possible to determine the ac-dc differ- ence to a precision considerably better than the fluctuations in the source, and the procedure described tends

to eliminate drifts in both the test and standard instruments. If both instruments inherently respond to the same values (rms for example), low-order harmonics up

to several per cent ordinarily cause the test to be in error by much less than 0.1 per cent. Electrodynamic Instruments

The electromagnetic force between two conductors

with reasonable geometry and currents is so small that

to obtain satisfactory torque the effect is magnified by

coiling the conductors. The torque-weight ratios thus

obtained are still considerably less than those enjoyed

by dc instruments with permanent magnets, but rea-

* A. H. M. Arnold, "Alternating-current instrument testing

equipment," Proc. lEE, vol. 101, pp. 121-133; 1954.

' F. L. Hermach, "Power supplies for 60-cycle tests of instru- ments and meters," Proc. ISA, vol. 11, paper 56-21-3; 1956.

sonably rugged and portable pointer-and-scale instruments of the 0.1 per cent accuracy class, based on

Weston's designs, have been available for a number of decades. For increased resolution and accuracy, recently

developed commercial instruments, following Silsbee's

composite-coil design,® now use a taut-suspension sys- tem with the moving coil of a second instrument rigidly

fastened to it to provide an electromagnetic restoring

torque.''"^ This torque can be precisely determined by

measuring the current through the second instrument

with a resistor and dc potentiometer.

As the torque equations indicate, an electrodynamic

instrument responds to the square of the current, the

scale being marked to indicate current directly. As an unshunted ammeter (limited to about 0.1 ampere be- cause of the springs or ligaments which must carry the

current to the moving coil), it will be useful as a transfer

instrument over the range of frequencies for which the

average torque, T =/(/), is the same as with direct current. The chief factors which affect this in an actual

instrument are: 1) capacitances between turns and between coils, which alter the currents; 2) eddy currents

in neighboring metals, which alter the field that links

with the moving coil; and 3) electrostatic torque, caused

by differences in potential between the fixed and moving coils. In general, these set an upper limit of about 1 kc for such instruments. The frequency ranges of ammeters, which use shunts across the moving coils, and

voltmeters, which use series resistors to limit the current, are more sharply limited by the inductances of the

coils. The manufacturer must insure that the ratio of the

impedances of the two parts of the divided circuit of the

ammeter is equal to the ratio of resistances, to the full accuracy of the measurement over the frequency range

of interest. Similarly he must insure that the magnitude

of the impedance of the voltmeter (defined as the ratio

of the applied voltage to the current through the coils)

is equal to its dc resistance. By connecting resistors in series with the coils of the ammeter and capacitors

across portions of the voltmeter, the upper frequency

limit for 0.1 per cent error in commercial instruments can be raised from 100 cps to about 1000 cps, and for 0.25 per cent error to about 2500 cps.^"

Because the mutual inductance between the coils changes with scale position (the operating torque depends upon this) complete compensation cannot be at- tained at all scale positions. Mutual inductance intro-

' F. B. Silsbee, "Composite coil electrodynamic instruments,"

/. Res. NBS, vol. 8, pp. 217-264; 1932.

' G. F. Shotter and H. D. Hawkes, "A precision ac/dc comparator

for power and voltage measurements," Proc. lEE, vol. 93, pp. 314-

324; 1946.

* J. Sorge, "A new precision instrument for ac power measurements," VDE Fachberichte, vol. 17_, pp. 27-30; 1953.

' R. F. Estoppey, "Inductronic Electrodynamometer," Confer- ence on Electronic Standards and Measurements, paper 35; August,

1958.

J. H. Miller, "Frequency compensation of ac instruments,"

Trans. AIEE, vol. 70, pp. 217-221; 1951.

18
238 IRE TRANSACTIONS ON INSTRUMENTATION December

duces other errors as well, so that the electrodynamic

ac-dc transfer instruments designed and constructed at the U.S. National Bureau of Standards^^ and the National Physical Laboratory of South Africa^^^ are nor- mally operated only over a narrow range of deflections

about the position of zero mutual inductance, by the use of a continuously adjustable external multiplier for the first instrument and a torsion-head for the second.

In the NBS instrument (see Fig. 1), two sets of coils are

astatically arranged to eliminate the effect of uniform

external fields, and a strip suspension and light-beam

pointer with a scale 2 meters from the instrument insure

definite readings and permit a scale factor at 0.01 per

cent/mm at currents from 0.1 to 20 amperes. The instrument is normally used as a wattmeter or ammeter, and

a similar but older companion instrument as a voltmeter

(10 to 600 volts).^' The NPLSA instrument is quite similar, but may be connected as a voltmeter as well (0.05 to 5 amperes and 25 to 500 volts). The ac-dc difference

of each of these instruments is believed to be known to better than 0.01 per cent at power frequencies and to perhaps 0.1 per cent up to 3 kc (above 50 volts), verified

by careful study and by comparison with electrothermic

and electrostatic instruments.

Electrostatic Instruments

The electrostatic force between two conductors at reasonable spacings is very low at normal line voltages.

The resultant torque can be multiplied, as in the Kelvin

electrostatic voltmeter, by interleaving the fixed and

moving conductors (plates), but unfortunately only on a one-to-one basis, so that torque-weight ratio remains

low. Thus electrostatic instruments, as distinct from

electrodynamic instruments where the fixed coil may be

designed to have more ampere-turns than the moving

coil, seem rather delicate except at voltages above about

1 kv, and are rarely available commercially for precision

measurements. However, taut-suspension, light-beampointer constructions, using improved optical systems

and modern materials, have begun to appear commercially in Europe and may presage a renaissance of this

type of instrument.

Electrostatic instruments inherently respond to volt- age but may be adapted for current measurements by

measuring the voltage across a resistor carrying the

current. However, the voltage must be rather high (50

to 100 volts) for adequate accuracy and the shunt must

then dissipate considerable power at high currents, or an instrument transformer must be used (either a current transformer with a resistor in its secondary, or a

voltage transformer between a resistor and the instrument).

" J. H. Park and A. B. Lewis, "Standard electrodynamic watt- meter and ac-dc transfer instrument," J. Res. NBS, vol. 25, pp. 545-

579; 1940.

J. W. Whittaker, "A precision electrodynamometer standard and

ac/dc transfer instrument," Proc. lEE, vol. 101, pp. 11-20; 1954. " F. K. Harris, "A suppressed zero electrodynamic voltmeter,"

J. Res. NBS, vol. 3, pp. 445^57; 1929.

Fig. 1 —NBS electrodynamic ac-dc transfer instrument

(external multiplier and scale not shown).

Electrostatic Instruments also have a number of sources of ac-dc differences. Among these are: 1) the

series impedance of the suspension or springs, 2) surface

charges on insulation in the electric field, which can modify the field with direct voltage applied, and 3) imperfectly conducting films on the plates which can cause

the field strength between the plates to differ on direct

and alternating voltage." None of these limit the fre- quency range as sharply as does the inductance of electrodynamic instruments, and electrostatic voltmeters may be accurate to perhaps 1 mc. However, when

an electrostatic instrument is used as an ammeter the

impedance of the 4-terminal resistor must be equal to

its dc resistance. (If a transformer is used, it is no longer an ac-dc transfer instrument.) This is difficult to achieve

above about 20 kc. Electrostatic voltmeters with ranges from 50 to 160

volts have been the principal ac instruments for over

40 years at the National Physical Laboratory in England" and somewhat modified instruments of the same

basic design are similarly used at the National Standards Laboratory in Australia. The ac-dc differences of the NPLE instruments are known to 0.01 per cent at power frequencies and to better than 0.05 per cent up to 100 kc (verified by calculations of the known sources of error and by comparison with electrodynamic and elec- trothermic instruments and with other electrostatic

" R. S. J. Spilsbury and A. Felton, "The electrostatic voltmeter

as a dc/ac transfer instrument," /. lEE, vol. 89, pp. 129-137;

1942.

19
1938 Hermach: AC-DC Transfer Instruments for Current and Voltage Measurements 239

instruments having noble-metal conductors). Special

transformers and resistors have been constructed and

tested to permit the measurement of current and of voltage below 20 volts, and resistance voltage dividers

have been evaluated for voltage measurements to 1000

volts (at audio frequencies).*

Electrothermic Instruments

Easily measurable effects are produced by the heating

of short straight metallic conductors or very small beads

of semiconductors, having very low electrical timeconstants. These effects can thus be independent of fre- quency up to 100 mc or more. In addition, the effects can be measured electrically, rather than mechanically,

with high precision and freedom from mechanical

resonances. These are the chief advantages of electro- thermic instruments.

Because their output is electrical, and because of their

low reactance, thermocouple instruments and bolometer

bridges are the only two forms now generally used. In commercial thermocouple instruments the temperature

rise of a conductor (heater) is measured with a thermocouple and millivoltmeter. The combination of the

heater and thermocouple is called a thermal converter. Many commercial bolometer bridges now contain bead

thermistors (temperature-sensitive resistors) and are unbalanced by the temperature rise caused by the heating of the applied current. The unbalance voltage is measured or balance is restored by additional currents

of a different frequency, which are measured. Thermocouple instruments of the ^ to 2 per cent class have long

been commercially available for current and voltage

measurements, while most commercial bolometers are used for power measurements (absorbing all of the

measured power). In either case undesired thermal effects, such as high ambient temperature coefficients,

accelerated aging of the heater at the elevated temperature, and transfer of heat to other parts of the measuring element have limited the sustained accuracy obtainable. However, by using such instruments solely as ac-dc

transfer instruments, calibrating them before and after each reading, these effects can largely be eliminated.

Studies at the National Bureau of Standards^^ have

shown that properly designed thermal converters may

be used as ac-dc transfer instruments at currents from 1 ma to 50 amperes with an accuracy approaching 0.01

per cent at power and audio frequencies and very probably up to 200 kc. Their low reactance also makes possi- ble transfer voltmeters from 0.2 to 600 volts (at 133

ohms per volt) with wire-wound series resistors that

have ac-dc differences less than 0.03 per cent at audio

frequencies. Voltmeter elements with film resistors in coaxial lines terminated by thermal converters have

been constructed and intercompared to 0.05 per cent to

^ F. L. Hermach, "Thermal converters as ac-dc transfer standards for current and voltage measurements at audio frequencies,"

J. Res. NBS, vol. 48, pp. 121-138; February, 1952.

10 mc.^^ The calculated ac-dc differences of the converters and of the voltmeter elements were verified by intercomparisons of different ranges and by comparisons

(over limited ranges) with electrodynamic and electrostatic instruments. A Lindeck deflection^ potentiometer

is used at NBS with these converters and voltmeter ele- ments to provide high resolution (0.005 per cent/mm at rated output emf) and rapid reading.

Similarly it has been shown that a properly selected

and mounted pair of indirectly heated thermistors may

be used in a bridge as an ac-dc transfer instrument for an audio-frequency potentiometer to better than 0.02

per cent at frequencies from 0.2 to 20,000 cps." Thermistors are also used for ac measurements at the Physikalisch-Technische Bundesanstalt in Germany. Their

rather long thermal time constants (about 10 seconds)

provide good low-frequency response but make highly

stable sources necessary.

The small ac-dc differences of thermal converters and

thermistor bridges at audio frequencies are caused al- most solely by: 1) thermoelectric effects in the heater,

which affect the temperature rise on direct current, but

because of thermal inertia do not affect it on alternating

current; 2) integration errors at low frequencies, which

are inversely proportional to the square of the fre- quency; and 3) small reactance effects in the voltmeter

elements, such as capacitance currents between parts of the multiplier, and from the multiplier to its surrounding shield. Fortunately, it is possible to obtain thermal

converters with heaters of manganin or other alloys of low thermoelectric effects and of sufficient length to make the ac-dc differences 0.01 per cent or less from 20

cps to 200 kc. A single 10-ma thermal converter is used with multirange shunts and series resistors for ac measurements

in the U.S.S.R., both in a central standardizing laboratory and in convenient consoles at branch laboratories.^' Series and shunt resistors are also used in the

laboratories of the Germany Authority for Weights and

Measures, but with two converters in a differential

circuit.^" As a further modification of the transfer principle, the heater of the thermal converter alone (without

the shunt or series resistors) can be switched to direct current for the dc calibration. Only a small dc source is

1' F. L. Hermach, "Electrothermic instruments for the measure- ment of alternating current and voltage," Proc. NPL Symposium on

Precision Electrical Measurements, HMS Stationary Office, London,

Eng., paper No. 15; 1955. " F. C. Widdis, "The indirectly heated thermistor as a precise ac-dc transfer device," Proc. lEE, vol. 103, pt. B, pp. 693-703; 1956.

1' H. J. Schrader, "A-C potentiometer measurements with in- directly heated NTC resistors," Electrotech. Z., vol. 73-A, pp. 547-

549; 1952.

1' K. P. Shirokov, "An installation for calibrating ammeters and

voltmeters at high frequencies," Proc. Russ. Inst. Meteorology, vol. 24, p. 24, 1954. W. Rump, "On the exact absolute measurement of ac voltages

and a potentiometer for testing ac instruments," Elektrotechnik, vol.

5, pp. 64-67; 1951. F. L. Hermach and E. S. Williams, "Multirange audio-fre- quency thermocouple instruments of high accuracy," J. Res. NBS,

vol. 52, pp. 227-234; 1954.

20
240 IRE TRANSACTIONS ON INSTRUMENTATION December

required, making possible a convenient portable acces- sory to a dc potentiometer, which, in a recently improved model (see Fig. 2) provides ranges of 7.5 ma to 20 amperes and 0.5 to 600 volts with an accuracy of 0.05 per cent at frequencies from 5 to 50,000 cps. Other Instruments

A transfer instrument is an essential part of any ac- curate ac potentiometer. Such potentiometers have been

used to some extent in Europe but rarely in the United

States, except for special measurements. They require

phase as well as magnitude balance, and with the usual

tuned detector, measure only the fundamental component rather than the rms value. An electrodynamic

milliammeter is most commonly used with a standard

cell to standardize the current through the resistance

network, but a thermal converter or thermistor bridge

could serve equally well and over a wider range of fre- quencies. ^^'^^ Recent studies^^ have shown that some

converters have excellent long-time stability if they are maintained at a constant ambient temperature.

A method of determining the equality of a dc voltage

and the crest-to-crest value of an ac voltage to better

than 0.005 per cent has been developed at the National

Standards Laboratory of Australia^^ as well as a method

of generating an ac voltage wave with no harmonic

greater than 0.001 per cent of the fundamental. The

equipment has been used to verify the ac-dc performance of the NSL electrostatic voltmeter to better than

J. J. Hill, "A precision thermo-electric wattmeter for power and

audio frequencies," Proc. lEE, vol. 105, pt. B, pp. 61-68; January,

1958.^ W. E. Smith and W. K. Clothier, "Determination of the dc/ac

transfer error of an electrostatic voltmeter," Proc. lEE, vol. 101, pp. 465-469; 1954.

Fig. 2—Volt-ampere converter for ac current and voltage measurements.

0.01 per cent at 50 cps and may be considered to be a

crest-value ac-dc transfer instrument.

High-gain amplifiers with a rectifier in a feedback

loop, and driven rectifiers based on the ring modulator,

may be used with a potentiometer to measure the aver- age value of an ac voltage wave to 0.1 per cent or better,

provided the galvanometer or other detector properly

integrates the rectified output. Their accuracy must also be verified by the procedure outlined for ac-dc transfer

instruments. At present this is most easily done by comparison with rms instruments in circuits with voltages

of known waveforms.

Reprinted from IRE TRANSACTIONS

ON INSTRUMENTATION

Volume 1-7, Numbers 3 & 4, December, 1958 PRINTED IN THE U.S.A.

21
Section 4

NBSIR 84-2903

AN INVESTIGATION OF THE

UNCERTAINTIES OF THE NBS THERMAL

VOLTAGE AND CURRENT CONVERTERS

F. L. Hermach

U.S. DEPARTMENT OF COMMERCE

National Bureau of Standards

Center for Basic Standards

Electricity Division

Gaithersburg, MD 20899

April 1985

Final Report, NBS Contract NB81 SBCA071

1

Submitted February 28, 1983

Revised August 1 984

This report and the work described in

it were supported by the Calibration

Coordination Group of the U.S.

Department of Defense.

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige. Secretary

NATIONAL BUREAU OF STANDARDS. EnMst Ambl«r. Director

22
TABLE OF CONTENTS

Page

Summary 1

I. INTRODUCTION 3

A. Scope of Contract and Reports 3

B. Present NBS Standards 3

C. Ranges and Uncertainties of Calibrations 4

II. SYNOPSIS OF THE AC-DC STUDY 5

III. MJTC's AND COMPARATOR 7

A. Comparator 7

B. Comparison of MJTC's 10

IV. REFERENCE TE's FOR F^ TVC's TO 10 kHz 12

V. EXTENSION OF FREQUENCY RANGE 13

A. Thermoelements 13

B. Thermal Voltage Converters 15

VI. EXTENSION OF CURRENT RANGE 16

VII. EXTENSION OF VOLTAGE RANGE 19

VIII. WORKING STANDARDS 23

A. Current Standards 23

B. Voltage Standards 23

IX. THERMOELEMENT REPLACEMENT 24

X. ERROR ANALYSIS - GENERAL REMARKS 25

XI. RANDOM (STATISTICALLY EVALUATED) ERRORS 29

A. MJTC Comparator 29

B. Other Comparators 30

C. Statistical Parameters and Tests 30

D. Pooled Standard Deviations 31

E. Experimental Designs and Tests 31

XII. ESTIMATES OF RESIDUAL SYSTEMATIC ERRORS (Not Statistically

Evaluated) 32

XIII. COMBINATION OF UNCERTAINTIES 33

XIV. CONCLUSIONS 35

A. Uncertainties of the NBS Standards 35

B. Calibration Uncertainties 36

XV. RECOMMENDATIONS 37

A. Maintenance of AC-DC Standards 37

B. Quality Control of the Calibration Process 38

C. Other Recommendations 39

23
Table of Contents (Continued)

Page

XVI. ACKNOWLEDGEMENTS 40

REFERENCES 41

TABLES:

1 Comparison of Four MJTC's 43

2 Differences Between 1981 & 1976 Evaluations of MJTC's .... 44

3 Results of Comparisons of FX and FY with MJTC's 45

4 Results of TVC Tests with Twin Resistors 46

5 Ampere-Range Comparisons 47

6 AC-DC Differences of 1A to 20A TE's 48

T Results of Evaluation of FXg and FYg 49

8 Comparisons of F^tvC's 50

9 AC-DC Differences of F^ TVC's with FX2 and FY2 51

10 Comparisons of Working and Reference Standard TE's 52

11 AC-DC Differences of Working Standard TE's 53

12 Comparisons of Fj TVC's 54

13 AC-DC Differences of Fy TVC's 55

14 Pooled Standard Deviations, S^,, 56

15 Estimated Limits of Residual Systematic Uncertainties .... 57

16 Calculation of Uncertainties 61

17 Summary of Uncertainties of NBS AC-DC Transfer Standards . . 64

18 Comparison of Estimated Calibration Uncertainties with

Present Allowances 65

19 Estimated Uncertainties for Special Tests 66

24
Table of Contents (Continued)

FIGURES: Page

1 Road Map 67

2 Comparisons of MJTC's 68

3 50 mA to 5 mA Comparisons 69

M 50 mA to 1A Comparisons 70

5 Comparator with n-Compensation 71

6 Diagram of EMF Comparator 72

APPENDICES:

1 n-Corapensation and n-Measurements 1-1

2 EMI, Harmonics and DC Offset 2-1

3 Effect of Relay Off-Time 3-1

H Effects of Bead Heating 4-1

5 Impedances and Substitution Corrections 5-1

6 Recommendation of the Working Group on the Statement

of Uncertainties presented to Comite International

des Poids et Mesures 6-1

25
AN INVESTIGATION OF THE UNCERTAINTIES OF THE

NBS THERMAL VOLTAGE AND CURRENT CONVERTERS

SUMMARY

This investigation required a complete redetermination of the ac-dc

differences of the NBS ac-dc transfer standards, from 20 Hz to 100 kHz. It

involved the following steps:

1 . Setting up a new and considerably more accurate set of primary

ac-dc standards, based on multi junction thermal converters (MJTC's) and a precise eraf comparator for comparing pairs of standards,

2. Using them to evaluate 5-mA single junction thermoelements (TE's)

and a 10-V thermal voltage converter (TVC), up to 10 kHz,

3. Extending the frequency range by complex "bootstrap" techniques

that were developed for this, and justifying these techniques,

4. Extending the current and voltage ranges of sets of

reference-standard TE's and TVC's by appropriate build-up methods, up to 20 A

and from 1 to 500 V,

5. Using these reference standards to evaluate the NBS working

standards that serve for calibrating clients' standards,

6. Determining the random errors of each of the above steps,

7. Studying and evaluating residual systematic errors (after known

corrections were applied) by special tests and experimental designs, then

estimating their individual limits (magnitudes).

26
8. Combining these estimates and random errors in a logical way to

arrive at estimates of the uncertainties of each of the many reference and

working standards (following BIPM Committee guidelines),

9. Carrying out a number of largely mathematical studies relating to

the investigation and these sources of ac-dc differences, and, finally,

10. Recommending steps to be taken to improve and Insure the continued

accuracy of the NBS calibration program for ac-dc differences.

As a result of this work it is believed that the accuracy of the NBS

standards and the output of its calibration process for ac-dc transfer are on

a much firmer basis than heretofore, with much better documentation.

The studies showed that in certain ranges of voltage, current and

frequency considerable improvement in the calibration accuracy can be

attained — factors of 2 to 5 for high-quality single-range TE's and TVC's.

Considerable improvement should also be possible for commercial multlrange

instruments after further study of their stability and other characteristics.

For very special tests, such as international comparisons, accuracies

approaching 1 ppm should be possible.

27
I. INTRODUCTION

A. Scope of Contract and Reports

In brief, this investigation, performed under contract with NBS,

required a theoretical and experimental analysis of the uncertainties in

maintaining and using the NBS ac-dc transfer standards for current and

voltage measurements at frequencies up to 50 kHz, plus suitable documentation

of the study. In effect, the contract required a redetermination of the

NBS ac-dc transfer standards. This was carried out to 100 kHz for voltage

measurements because of frequent calibration requests at the higher

frequency.

The contract was completed and the final report was submitted to NBS in

February 1983. The report was revised and the appendices were added in

August 1984 for publication as an NBS Internal Report.

B. Present NBS Standards

The basic or primary ac-dc transfer standards at NBS at these

frequencies are thermal current and voltage converters, described in [1] and

[2]. The working standards used in the calibration of client thermal voltage

converters (TVC's) are described in [3]. Papers [2] and C3] are appended to

this report for ready reference.

Extensive research was carried out at NBS to establish these standards

and verify their accuracy. The basic principles and brief descriptions of

the investigations and of the many special tests required for these

pioneering developments are given in the references. The work described in

[1] (1952) established the ac-dc transfer to 0.01 51 (100 ppm) for current

measurements from 1 mA to 20 A and voltage measurements from 0.2 to 750 V at

audio frequencies. In the work described in [2] (1966) a basic set of 5 to

20-mA 3 ingle-j unction thermoelements (TE's) was evaluated to a few ppm, and

28
two selected TE*s were used with special series resistors as thermal voltage

converters up to 500 V to 10 ppm. An emf comparator was developed to compare

the ac-dc differences of pairs of converters to ±2 ppm.

Fifteen comparisons of adjacent or similar ranges are required at each

frequency with this primary set of TVC's (called F^), to verify the build-up

(step-up) from 0.5 to 500V. In [33, Williams described a set of TVC's which

required only 7 comparisons, thus considerably shortening the time required

for routine verification of accuracy. His set (called Fy) is now the working

standard for the determination of ac-dc differences of TVC's sent in to NBS

for calibration. In a later paper [6] he described a similar set of shunts

with two thermoelements (TE's) for evaluating thermal current converters

(TCC's). Because of grounding and capacitance difficulties with the shunts,

however, TE's are still used as the working as well as the reference

standards for TCC calibrations. The contact-type TE's formerly used at 1 A

and above were replaced with insulating-bead types, to eliminate the

troublesome direct coupling between the heater and thermocouple circuits.

The reference TE's from 25 to 500 mA were also made to NBS special

specifications a number of years ago. Each has a Evanohm^ heater, with

Evanohm leads from it to the glass press, to minimize ac-dc differences from

Thomson and Peltier effects, and from the skin effect of the magnetic leads

normally used.

C. Ranges and Uncertainties of Calibrations

The following table shows (for ready reference) the stated uncertainties

actually available in the calibration of client ac-dc converters at the start

of this Investigation. It is considered to be very unlikely that the errors

would exceed these values.

Certain commercial products or materials are identified in this paper to

specify the experimental procedure adequately. Such identification does not

imply recommendation or endorsement by NBS, nor does it imply that they are necessarily the best available for the purpose.

29
Frequency

2-5

Hz

5-20

Hz

20-20k

Hz

20-50

kHz

50-100 0.1-0.5

kHz MHz

0.5-1

MHz

Voltage limits (V) 50 100 1000^ 1000 600 1 00 100

Current limits (A) 0,05 0.05 202 15

- Uncertainty (percent)

3

Multirange TVCs 0.02 0.01 0.005 0.007 0.01 0.02 0.03

Coax single range

TVCs

0.02 0.01 0.002 0.003 0.005 0.02 0.03

TCC (.005 to 5 A) 0.02 0.01 0.005 0.01

TCC (>5 to 20 A) 0.01 0.02

200 volts at 20 Hz, increasing to 1000 V at 200 Hz.

^5 amps at 20 Hz, increasing to 20 A at 200 Hz.

^The lower uncertainty applies at the crossover frequencies. Uncertainties

may be increased if the ac-dc differences are large or affected by selfheating.

The larger uncertainties of the multirange converters reflect, in part,

the poorer precision and stability of these more complex but very convenient

instruments

.

II. SYHOPSIS OF THE AC-DC STUDY

Quite a number of years have elapsed since the comparisons were made

which formed the basis for the present accuracy estimates. UBS now has new

types of TE's. A series of comparisons of the NBS current converters was started a few years ago, but has not been completed. For these reasons a meaningful study of uncertainties required setting up a new base (primary)

group of TE's, as well as carrying out new complete series of interrange

comparisons of both the current and voltage standards (reference and working)

for extending the ranges from the base (build-up).

30
The same emf comparator and the same or new types of TE's could have

been used for the base and for these comparisons, as before. However, It was very desirable to use, Instead, a group of the newer multl Junction thermal

converters (MJTC's) and a new comparator, described In [4], (1976), which Is

also attached to this report. At that time the ac-dc differences (d'a) of

these 5- to 12-V standards were evaluated to 0.5 ppm up to 10 kHz for the NBS

Absolute-Volt Program. However, the converters have not been used since

then. For this study the apparatus was set up again with the MJTC's to form

an excellent base for directly evaluating the corresponding ranges of the

reference set of thermal voltage converters (TVC's), and the TE's used In

them as current converters. A considerably better accuracy was thereby

attained. Thus this study was based on the new MJTC's as primary standards,

with subsequent experimental comparisons with TE's, which were then used for

the extensions in frequency and range.

Special studies and tests were made to extend the frequency range from

10 to 100 kHz for voltage measurements and to 50 kHz for current measurements. The d's of the 5-mA reference TE's from all known causes except bead

heating should theoretically be independent of frequency from 1 kHz up to at

least 100 kHz. To buttress the theory they were compared with other TE's of

different types and construction, which should also be independent of frequency in this range. As described later in this report, the agreement was excellent.

Resistors of calculable reactance are used with a pair of the reference

TE's to form the voltage ranges of the F^ reference set. In the 1966 evaluation [2], agreement of the 5 to 50 V ranges, where the computed reactance

errors were negligible, formed the basis for assigning d's. Since then evidence of dielectric loss effects has been found. To evaluate them the 10-V

range was compared with TVC's containing resistors of measured ac-dc characteristics, and with two of the NBS 30-MHz TVC's [5]. These rf TVC's have

carbon-film resistors, with agreement in d's to 0.156 out to 40 MHz. The

results showed that there was Indeed an ac-dc difference in the 10-V re- sistor, but that it was less than 6 ppm, even at 100 kHz.

31
A block diagram of these major steps in the evaluation of the NBS ac-dc

reference and working standards is shovm in figure 1

.

Analyses and tests were also made of the important problem of accurately

deducing the d's of the many ranges of the and TVC sets when the TE's

are replaced, without retesting all of the ranges. Many other analyses were carried out to evaluate systematic errors from known sources in TE's and in

the comparison of TE's. Suitable statistical analyses of random errors were

also carried out, and some additional sources of systematic error were

discovered.

III. MJTC*s AND COMPARATOR

A. CoBparator

For this study the precision MJTC comparator and associated equipment

were set up in the NBS AC-DC Laboratory, room Al4i|, MET, which has excellent

temperature stability, with a typical range throughout the day of ±0.6 ®C. A

simplified diagram of the comparison circuits and a description of the

procedure for comparing two MJTC's are given in [1]. Briefly, with a resistance divider a fraction of the higher emf of one of the MJTC's is opposed to

the full emf of the other. The small differences are read on a nanovoltmeter

(nVM), with ac, dc direct, dc reversed, and ac, successively applied to the

two heaters at equal time intervals (to minimize the effects of drifts).

From these the difference between the two d's is calculated.

Several important changes and improvements were made so that the

comparator could also be used for TE's. Brief descriptions follow.

(1) In place of oil immersion for the standards, the metal tank was thermally lagged. This was found to be adequate. Oil immersion would have

been impracticable for many of the NBS standards.

32
(2) When the two TE'3 have equal n's, (n - lAE/EAI, where I Is the

heater current and E the output emf), the steady-state balance is Independent

of small changes in the ac and dc supplies, greatly reducing the uncertainty

In the measurement and the need to keep the emf of one of the TE's constant

during a determination. However, n may range from 1.6 to 2.0 for TE's, so that some of these advantages are lost. A simple method of compensating the

comparator for different n's was developed. It Is described In appendix 1.

With this modification the basic equation for comparing two TE's is

A = d.j. - dg = (N(j - lig^)/nQE^f where T refers to the TE with the higher n, to

which the Llndeck potentiometer shown in figure 6 is connected, and S to the

other. and are the average readings of the nVM with dc and ac applied

to the heaters, respectively, and is the lowest emf of the two TE's. With

N in nV and E in mV, A is in ppm.

(3) An operational-amplifier integrator and timer (called an "averager") was designed and built to integrate the 1-V output of the nVM for

10 3, and display the average on a DVM. This significantly reduced the

uncertainty from Johnson and nVM noise, and greatly aided the operator.

Typically, repeated 10-s averages fluctuated by less than 1/10th of the

peak-to-peak noise seen in 10 s on the nVM.

The 11-kJ2 input resistance of the 3-stage divider was reduced to

1 kfl, by switching out the first stage when two single-junction TE's, with

their low output resistance, were compared. This improved the performance of

the nVM and reduced the Johnson noise of the circuit. The peak-to-peak noise

in 10 s was about 15 nV, about 3.5 times the calculated Johnson noise. This

is 1.5 ppm of the typical 10 mV rated output of a TE, and 5 ppm with the 3-mV

output encountered in Interrange comparisons.

(5) Adequate shielding and grounding are very important. All ground

leads were connected to the grounded metal tank in which the TE's were

placed. A bypass capacitor was connected between each lead and the tank, to

reduce any lead-borne interference. Although there was no direct evidence

33
that It was significant, electromagnetic interference (EMI) can be

particularly troublesome at TV frequencies, where leads are near a quarter

wavelength and so pick up electromagnetic energy efficiently.

A simple direct test was devised to see if EMI caused significant

errors. With the TE's and supplies connected, but with all power supply

switches off, a "phantom" ac-dc test is performed, using the formula for A

given in IIIA(2) above with the value of ngEji^ to be used in the actual test.

The result should be zero within the random-error limits. As shown in

appendix 2, this test is sensitive enough to detect significant errors.

(6) If there are ac components in the input, chopper-type dc

electronic nanovoltmeters (such as the one normally used) may be subject to

intermodulation effects which can result in false zero-order (dc) outputs.

These normal-mode errors were noted in the MJTC study [4], and were reduced

by filtering. Careful measurements showed that at the commonly-used

frequencies the ac output of a TE is typically less than 1 yV, and that this

is too small to cause a significant error from this insiduous effect.

(7) About midway in this study, a zener-diode protective circuit,

discussed in section VII, was installed at the input to the TVC's, to reduce

the possibility of burnout from over-voltages.

A detailed diagram of the comparator, with these changes, is shown in

figure 6. It is quite similar to the comparator shown in figure 8 of [3],

attached, and the same precautions in its construction, described in [3],

were taken. The first stage of the divider is set at zero when the Read 2

position of the switch is used. Shielding, polarity markings, and the fine

controls of the Lindeck potentiometer, are not shown. Correct polarity is

assured if the nVm goes upscale (more positive) with the Set key down when

the input to the TVC's or TE^s is increased.

The conventional definition of ac-dc difference of a TE is

d = (Ig-I^)/I^, where 1^ is the arithmetic average of the two directions of

the applied dc current required to give the same output emf as the rms

34
current, I^. From this » l^il + d), so that an unknown ac current can be

measured directly, If d is known, by observing the emf with I^^ applied and

measuring the two polarities of dc current which give the same emf. The same formulas apply for TVC's, with V replacing I.

In a. recent paper, however, [7] Inglls has shown that this definition is

in error if the TE has a proportional dc reversal difference, p - |l^-l2|/I(j,

with E^-E2» where the subscripts, 1 and 2, refer to the two polarities. He

has proposed a new definition, d^-dg^-I^)/:^, where I^j-I^-Ij and

(E-|+E2)/2"Eg. It is free of this error, but can be awkward to use directly,

for ac measurements. The difference between the two definitions is

dependent upon a number of factors. However, for a square-law TE in which p

is well behaved (approximately Independent of E), the difference between the

two definitions is only about 3p^/8, If p = 5x10"^ (rather large for a precision TE Intended for ac-dc measurements) the difference is less than

0.1 ppm.

At NBS the primary standards are the MJTC's, with p's much less than

1x10~^. The other TE's are evaluated by the steps shown in figure 1, with

the conventional definition of d.

Additional analysis shows that, with the conventional definition, there

are small error terms which should be added to the basic formula for the NBS

emf comparators when TE's having large dc reversal differences are compared.

However, they are about the same magnitude as the 3p^/8 term just given, and are therefore also negligible for the NBS TE's, which have small and well

behaved p's.

B. CoBparlson of MJTC*s

The ac-dc difference of 4 MJTC's were carefully compared at 6 to 10

volts (limited by voltage ratings) at 30 Hz, 1 kHz and 10 kHz. These are 4 of the 8 that formed the MJTC base described in [4]. The 4 are of different

design and manufacture. This diversity greatly strengthens the confidence in

35
standards, because sources of systematic error are more likely to be

discovered in such comparisons than by simply repeating measurements on one standard, or comparing like standards.

The results of the evaluations (including repeat tests on different

days) are given in table 1, as d^-dg in ppm for the MJTC's identified as T

and S. The supplementary designations of the MJTC's shown in parenthesis in

the figure, are those given in the 1976 paper.

The pooled standard error, s^, (standard deviation of the average) for

these measurements was 0.16 ppm. If the difference between two averages in

repeat tests is less than /2tS3 (t is Student's t) we have no evidence (at a given confidence level) that the difference is statistically significant. At

a 95% confidence level for k(n-1) degrees of freedom with k-19 (the number of

averages in table 1), and n=«4 (the number of determinations in each),

i/Tts^ » 1.4 X 2.0 X 0.16 » 0.45 ppm. None of the differences between

averages was as large as this.

The results of these measurements at each of the 3 frequencies are shown

more vividly in figure 2. Here the number near the head of each arrow is

d^-dg, where T is the MJTC at the arrowhead. The best estimate for

'^Gr25-1 ~^G44 = M is obtained from the paths between them, giving double

weight to the direct path. The best estimate for ^q^q is obtained by

assigning to each pertinent arrow half of the discrepancy between M and the sum of the two arrows involving G5C. Similarly dj^pj^^^ is determined. (NPL14

could not be used at 10 kHz because of its magnetic lead wires.)

The value assigned to each MJTC, on the assumption that the average of

all 4 is zero, is shown to the nearest 0.1 ppm in each circle. G10C was not

included in this group because it was compared only with G5C, in early

measurements, which were not repeated.

36
The same MJTC's were evaluated in 1976 [4] as part of a larger group.

Table 2 shows the differences between the present assignments and those of

1976. They are gratifyingly small, indicating the high stability as well as

the low ac-dc differences of these thermal converters of different design and

construction.

IV. REFERENCE TE»3 FOR F^ TVC'a TO 10 kHz

The first comparisons of the 10-V reference TVC with one of the MJTC's,

and subsequent intercomparisons of TE's, disclosed several problems. The TVC

was unstable because of variable contact resistance in its sliding

connectors. Some TE's had a peculiar dependence of ac-dc difference upon the

ac-dc sequence used, differing by as much as 3 ppm with ac, dc+, dc-, ac applied than with ac, dc-, dc+, ac applied. Several of the TE's required 15

to 20 s to come within 1 ppm of the final value after the input was switched, even though the off-time of the relay was less than 0.5 ms.

To overcome these problems simple modifications were made in the TVC

connectors and new 5-mA TE's were chosen after extensive screening tests.

Each consisted of two special TE's with Evanohm heaters, welded to the copper

skin of the CuNi lead-in wires to reduce thermoelectric errors.^ The two

heaters were connected in series, as were the two thermocouples, to obtain a rated output of 12 mV. They were temperature compensated by the method used

by Williams [3].

Two of them, labeled FX and FY, were compared as both current and

voltage converters with two of the MJTC's at 30 Hz, and 1 and 10 kHz, using

the 5- and 10-V F^ resistors for the TVC tests. The results are shown in

table 3. The third TE, FZ, was compared with FX to 100 kHz.

Later analysis indicated that welding to the copper instead of the core might

not greatly reduce the Peltier error.

37
The measured d's were gratlfyingly small. There was no appreciable d

from Thomson and Peltier effects in the heaters of any of the three TE's.

There were small low-frequency errors at 30 Hz, and evidence of impedance

errors (about 1 ppm at 10 kHz) in the TVC resistors.

To check on possible current-dependent d's at high frequencies, FX and

FY were paralleled and compared with FZ at 5 mA, with excellent agreement.

Puzzling drifts of some single-Junction TE's led to a study of transients caused by switching the input voltage during an ac-dc determination.

As shown in appendix 3, if the response of a TE to a step-change in input is a simple exponential with a time constant, t^, and the relay off-time in

transit is t^, (tQ<<tQ) then the proportional change in output emf at a time,

t, after switching (caused by the off time) is p - tQ/t^e^, where a » t/t^.

For the high-speed miniature relays now used, t^ is only 0.^ ms. For a TE

with a t^^ of 2 s (fairly typical) a time t-10 s is required to make p<1 ppm.

This is comfortably below the 30 s interval (time before reading nVm) now

used.

The present relays can be used only up to 200V in the circuit, and

high-voltage relays have much longer off times. Longer intervals might then

be required. Surprisingly, for t^ = ilx10~^ s, t-15 s, only a modest in- crease. Thus the excessively long drifts of some TE's are evidence of other

problems, of unknown causes.

V. EXrENSIOM OF FREQUENCY RANCX

A . Thermoelements

Because of reactances in the complex structure of the 5- and 10-mA

MJTC's, these direct comparisons could not be extended reliably in frequency.

The simpler TE's, with their smaller reactance errors, should have wider

frequency ranges. To verify this, several 5-mA TE's of different

construction and/or manufacture were compared with FX and FY as current

converters from 30 Hz to 50 kHz. Two of them were in the original 1966

38
primary set. All of them have Evanohm heaters and should have low

thermoelectric errors. These were determined from the 1-kHz MJTC tests, and were independent of frequency. The important criterion is that the measured

differences between each pair of TE's should be the same at higher

frequencies, giving strong evidence that reactance and other errors were negligible.

Unfortunately the measurements showed no such agreement. At 50 kHz

discrepancies of up to 4 ppm from the 1-kHz values were observed. It seemed

likely that some of the differences could be caused by leakage current

through the insulating bead which fastens the hot Junction of the

thermocouple of a TE to the center of the heater. [1,2] The bead resistance

is a complex function of frequency and of bead temperature and voltage. This

was verified by making additional measurements with added resistance between

the heaters of the two TE's, which changed the bead voltage of the ungrounded

one. (The thermocouple circuit is grounded at the emf comparator.)

Apparently the bead resistance of these TE's was less at 50 kHz than at

1.6 kHz, where bridge measurements were made earlier (as described in [2]).

Detailed analysis of bead resistance errors, given in appendix led to some surprising conclusions. They showed (1) that bead heating (caused by

leakage current from the heater to the thermocouple through the bead) is

twice as effective in raising the thermocouple temperature as heating from

the heater wire, (2) that if one end of the heater is grounded, no error would be caused by bead heating, and (3) this would be true even if a pair of

equal TE's, with heaters in series, is used, as in FX and FY. (In the

comparisons with the MJTC's described earlier, FX and FY were at the ground

end.)

With this background, the comparisons of pairs of TE's were made by

following a method used by Williams. They were compared as TVC's with two

nearly identical 30-V series resistors. One end of each of the heaters of

the two TE's under test could then be at ground. Appendix 5C shows that by

repeating the measurements with the resistors interchanged and averaging the

39
results, errors from small differences In the reactances of the resistors

could be largely eliminated, and the TE's could be evaluated as current

converters.

The results of these measurements are shown in table 4. Except at

30 Hz, there was no significant frequency dependence of the average of the 6

TE's up to 50 kHz, with FX as the reference. Thus there is good evidence

that the corrections assigned to FX and FY up to 10 kHz by comparisons with

the MJTC's could be extended without change to 50 kHz.

TE's are subject to low-frequency errors (1) caused by failure to

integrate the heating effect of the ac current wave correctly. For FX,

d = +0.9 ppm at 5 mA and 30 Hz. From these tests, 30 Hz values could be

assigned to all of the other TE's.

B. Theraal Voltage ConTerters

As shown in [2], the ac-dc difference, d^, of a TVC is dy = d^ + d^,

where d^ = (Z^-R^)/R^ and d^ = for the same output emf. In these

equations, is the magnitude of the ac impedance, defined as the ratio of

the input voltage to the current, I, through the ungrounded input of the TE

heater, is the dc resistance, and I^ is the dc current (average for the

two directions) required to give the same output emf as I^. Note that

includes the impedance of both the TE and the series resistor, and that the

equation for d^. is a perfectly general one.

Preliminary comparisons of the mid-ranges of the TVC's to evaluate high

frequency effects disclosed some disturbingly large differences at 50 and 100

kHz. Calculations showed that the d's caused by the reactances of the series

resistors should be much less than 1 ppm, and should vary as f . The

observed differences were roughly proportional to f, suggesting dielectric

losses as the cause. This supposition was strengthened by the -1 .0 ppm

average of F,10-FY and Fi5-FX in the comparisons at 10 kHz with the MJTC's

40
(table 3). The reactance errors should be completely negligible at this

frequency.

To investigate this further, F^io-FY was compared with a special TVC

that Haddad had studied in 1969 for his GWU Master's thesis under

CutkoskyCl 8] . He had determined the change in impedance (resistive and

reactive components) between 1.6 and 16 kHz. The comparisons indicated that

F^IO-FY had a Ad of about -1.5 ppm.

To carry this to higher frequencies, F-|io-FY and Fi20-FX were compared

with the corresponding ranges of the rf Model C TVC's, which had been

evaluated to MO MHzC5]. These TVC's have unspiraled carbon film resistors on cylindrical ceramic bases, as distinct from the spiraled metal-film

resistors, of the F^ set, with much smaller possibility of dielectric

effects.

The results of these several tests were not completely concordant, but

all of them pointed strongly to a small but significant d^ in FilO-FY. Later

tests showed that the dg of the resistor alone was about -5 ppm at 100 kHz.

VI. EXTENSION OF CURRENT RANGE

Range extension (build-up) is based on comparisons of adjacent ranges,

so that all are determined in terms of a base or reference range. Evaluating

the validity of the method is, of course. Important, and is often very difficult. For these current ranges, the extension was carried out by two

methods, as follows:

(a) From 5 to 1000 mA two almost identical TE's of one range were first

compared. They were then paralleled and compared with each of two TE's of

the next higher range, and so on. This is the method used in [2] to compare

the 5- to 50-mA TE's. Like that work, pairs of almost identical

low-reactance resistors were connected in series with the heaters to assure

that the paralleled currents were sufficiently in phase. The analysis of

41
appendix 5E shows that the phase angles of the two paths must be equal to

better than one mrad. This is readily measured on a commercial impedance

comparator, to 100 kHz. When the phase angles are negligible, the basic

formula for this method, taking into account small differences in the

characteristics of the paralleled TE's, is given in appendix II of [2]. This

method has the strong advantage that each TE is used at or near rated

current, so that the results are not affected by current-dependent errors.

(b) Above 1 A the heater resistances are too low to be measured on the

bridge, and mutual inductances can cause serious errors, so a more complex

approach following [1] was necessary. Adjacent -range TE's were compared at

the rated current of the lower range. To investigate current-dependent d's,

selected ranges were compared with shunted low-range TE's at frequencies low

enough so that the reactance errors of the shunts were negligible. Each such

comparison was made at two currents. All of the TE's had Evanohm heaters

with small thermoelectric effects independent of frequency, so that the

extrapolation of current-dependence could safely be made to 50 kHz.

The 5 to 50-mA build-up was made with the MJTC comparator in method (a).

It was convenient to start with a 50-mA (OF-19) MJTC of Wilkin's later, and

very different design[8]. His calculations showed that this MJTC should have

less than 1 ppm error up to 100 kHz. It agreed with the NPL-14 50-mA MJTC at

30 Hz and 1 kHz to 0.1 ppm, verifying low-frequency concordance. (NPL-14 is

frequency-limited because of skin effect in its magnetic leads.)

The results of these 5- to 50-mA comparisons are shown in figure 3. The

numbers near each arrow are d^-d3 in ppm, at the four frequencies shown in

the key, where the subscript T designates the TE at the arrowhead. The

columns headed "Average" are the averages of the two adjacent TE's, as determined by the two comparisons of that paralleled pair with the two TE's

above them in the figure. The numbers in each block are the d's determined

from the intercomparisons (starting with MJTC OF-19), with forced closure of

each triangle. (One third of the closure error is applied to each leg.) The

42
closure errors were very small, requiring adjustments of only 0.3 ppm or less.

The pooled standard deviation Sp^ of the average of four determinations

of d^-dg was only 0.28 ppm. These high precision tests showed excellent

performance in these Evanohm-heater TE's. At the low end of this rather long

chain of measurements, the values obtained for FZ and FY as current

converters agreed with those previously determined from the 5mA MJTC's to

1 ppm at 20 Hz^ and 50 kHz, and to 0.5 ppm at intermediate frequencies. Thus

systematic errors in the chain were very small indeed.

Of particular interest is the excellent agreement at 1 kHz (better than

0.6 ppm) between the MJTC's and the average of the two 50 mA and two 25 mA

TE's.^ Each of these four single-Junction converters has an Evanohm heater

and Evanohm stems (lead-in wires) from the heater to the glass press (which serves as a good heat sink). Thus the Peltier as well as Thomson errors of

each TE should be very small.. Any residuals would not likely be the same in

all of them, nor the same as those in the MJTC's. Thus the agreement

provides valuable additional confirmation that the ac-dc differences of the

NBS standards caused by thermoelectric effects are considerably less than

1 ppm.

The fast-acting relays controlling the inputs to the heaters limit the

high precision MJTC comparison equipment to 50 mA or 100 V (not simultaneously). For comparisons at higher voltages and currents, the 1966 com- parator [2] was used. The results of the paralleling measurements, method

(a), from 50 mA to 1 A are shown in figure 4, along with a comparison of one

of the 1-A TE's with FG-1 , a shunted 10-mA TE described In [6]. The build-up

is based on a 0 ppm assignment to the average of the two 50-mA TE's, together

with forced closure of each triangle of tests in the four loops up to 1 A.

'Extrapolated from the 30 Hz measurements.

At 1 kHz reactance errors and low frequency errors should be negligible.

43
For these measurements, Sp^^ was about 1.7 ppm, practically independent

of frequency. This is not as good as the precision of the MJTC comparator,

but is low enough to meet the needs for high-current calibrations. The

evaluation of FG-1 with the 1-A TE agreed with Williams' earlier tests [6] to

5 ppm, which is well within the limits of random error for the number of

stages involved.

The results of the comparisons of the 1- to 20-A TE's by method (b) are shown in table 5. These air-cooled TE's were made by the Weston Company a number of years ago to NBS specifications. Each has an Evanohm heater, with

an insulating bead at the hot Junction of the thermocouple. To guard against

current-dependent errors, some of them were compared with Williams' shunted

TE's. Each comparison was made at two currents, with no consistent evidence

of such dependence, so that the results of table 5 could be combined with the

values assigned to the 1-A TE to determine the d's of all of the ranges, as shown in table 6.

For these comparisons Sp^^ was about 1.7 ppm, again not greatly dependent

on frequency. The TE's showed rather large d's at 50 kHz, probably because

of skin effect in the lower ranges, and a peculiar "proximity effect" in the

5 to 20 A ranges. These TE's have heaters of C-shaped cross section rather

than tubular. For these ranges d was markedly dependent upon the position of

the return lead. [6]

YII. EXTENSION OF VOLTAGE RANGE

The extension from the base 10-V range of the F.| reference set of TVC's

up to the 1000-V and down to the 0.5-V ranges required 14 adjacent -range

comparisons at each frequency, using the FX and FY TE's with 13 plug-in

resistors [2]. Unfortunately, shortly after it was completed a major fault

in the ac amplifier burned out both of these excellent TE's. (A lower-range

TE connected in series with them as a fuse was more rugged.) Subsequently,

44
two unusual sources of error in the ac-dc comparisons were discovered,

casting some doubts on the accuracy of the TVC build-up.

The first of these was an unexpectedly large error caused by a do offset

voltage, Vqq, in the output of the same direct-coupled ac amplifier. Because

this was only one direction of dc, it would be integrated incorrectly if the

TE had a dc-reversal difference, g = AE/E, for the two directions of the same

dc voltage. As shown in appendix 2C, the error can be as much as e = gu/2,

where u = V^g/Vg^. Thus if u=1x10 and g = 5x10 , e can be 2.5 ppm. As

shown in 2B, this is much larger than the output error e' - q^e,^ caused by a small proportional error, e^, in measuring a small harmonic, V^, in an ac wave, V^, where q - V^/V^. If q - 1x10 and the error of an rms instrument

in measuring the harmonic is e^ » 5x1 0~^, the error in measuring the combined

wave is only 0.05 ppm.

The second source of error was an unexpectedly large skin effect in the

brass center conductor of the coaxial connectors. At 100 kHz, a 6-ppm error

was caused by an L-shaped connector sometimes used in front of one of the

0.5-V TE's in a comparison. The error decreased, of course, for higher

ranges with their higher resistances.

The effects of the dc offset and troublesome dc transients in the amplifier were eliminated by using a wide-range output transformer. The reference

plane in each TVC-comparison was taken as the center of a coaxial "T"

connector to which each of the two TVC's was thereafter directly connected.

Two new 5-mA TE's, FXg and FY2 were assembled to replace FX and FY.

They were evaluated by comparisons with MJTC G44 to 10 kHz and with FZ from

2.5 to 5 mA up to 100 kHz. F^10-FY2 was evaluated at 20 Hz, 1 and 10 kHz

with G4H as a TVC. Because of the importance of the frequency extension, the

tests described in section V-B were repeated with FY2. Results of these

extensive measurements are shown in table 7.

I

45
The results of the interrange comparisons of the F-| set are shown in

table 8, and the values assigned to each range, with d of F^io-FY2 from

table 7 as the basis, are shown in table 9. In the comparisons each TE is

used at two currents, with a series resistor of the given range. If its d^

is current dependent, a correction must be applied at each step. Since

dy-d^+dc (see section V-B), dy2-dyi --dQ2-d(ji , at the two currents

corresponding to subscripts 2 and 1 , on the assumption that d^ is not voltage

dependent.

Unfortunately each of these new TE's has a significant d^, which is also

somewhat current-dependent, as shown by the results in table 7. It was difficult to evaluate the current dependence accurately enough, because of

the low emf's at the low currents. Any inaccuracy is multiplied by the

number of steps from the 10-V base range to the extreme ranges. To reduce

the uncertainty, an average value of dc"^c2'^c1 calculated from all of

the values for both TE's from 1 to 100 kHz. Subscript 2 applies at 5 mA and

subscript 1 at either 3.3 or 2.5 mA. This value, +0.8 ppm, was applied

uniformly in calculating the d's in table 9 from the comparisons of table 8.

The tests with FZ showed that it was independent of frequency from 1 to

100 kHz. At 20 Hz the values of d^ were taken directly from table 7.

As a check on the accuracy of this average value for d^, the ac-dc

difference, dg, of the series resistor of each range was calculated at 1 kHz

by formulas given in appendix 5D, starting with the values of d^ of the TE's

as given in table 7, and d^ as shown at 0.6 V in table 9. At this frequency

each dg should be negligible. The computed values ranged from +0.4 to -0.9

ppm, and the average for all voltage ranges, with regard to sign, was only

-0.1 ppm. This is an excellent check indeed.

Both FXg and FY2 have d^'s which are positive, and so cannot be caused

by Thomson effects in their heaters [1]. Unfortunately they also have

unusually large negative dy^'s for their heater impedances (-10 and -12 ppm,

respectively), calculated from the measurements at 1 kHz. These values of d^

46
and are indicative of large Peltier effects at the heater to lead-in

Junctions.

At higher frequencies the even larger dy^ for the 0.6-V ranges (TE's

alone) are caused by skin effect, which increases the resistance of the

magnetic heater leads, causing positive ac-dc differences.

With the F., TVC set these factors require troublesome corrections, but

do not invalidate the results, as the correlation at 1 kHz shows. Indeed,

they provide excellent checks on the correctness of the basic build-up

formulas.

It is to be noted that these TE's cannot be used interchangeably; i.e.

F^10-FX2 is not known from table 9. Substitution corrections are discussed

in section IX.

The TVC comparisons below 200 V were made with the MJTC comparator.

Because the thermocouple emf's of the higher-range TE in each pair were low

(3 mV or less), the precision was poorer than with 10 mV outputs. The

pooled standard deviation, Sp^, was 0.52 ppm. The 1966 comparator was used

for the higher voltage ranges, with the original TE's, FX and FY. To avoid

repeating these measurements the substitution formula given in section IX was used to calculate the d^'s with FX2 and FY2.

A study of protective circuits to guard TVC's frcxn burnout, sparked by

the loss of FX and FY, showed that a simple combination of a pair of

back-to-back zener diodes directly in parallel with the TVC's and a 10-raA

expendable TE in series with the combination should provide adequate

protection against reasonable kinds of over voltages. The zener voltage

should be near, but not less than, twice the supply voltage, and there are

limits on the zener resistance and the time constant of the fuse (which is

why a TE was chosen). The method does work well, and TE's of large dc

reversal difference, which are relatively inexpensive, can serve well as fuses, but a CRO across the TVC's is advised, to monitor the wave form.

47
VIII. UORKING STANDARDS

A. Current Standards

The working standards which are used in calibrating thermal converters

sent to NBS were evaluated by direct comparison (heaters in series) with the

corresponding ranges of the reference converters, in the calibration console

normally used for such measurements. This had the advantage of giving direct

information on the precision of calibrating TE's as current converters

(TCC's). Usually the input line to the working standard was grounded. To

check on possible bead resistance errors, some measurements were repeated

with the reference standard at ground. (The ground should never be placed on

the line connecting the two heaters, because capacitance and leakage currents

from the ac and dc sources to ground could cause the two heater currents to

differ.)

The results of these comparisons are shown in table 10, and the values

assigned to the working standards are shown in table 11. The overall s„ was

2.4 ppm.

B. Voltage Standards

The Fy set of TVC's for calibrating TVC's sent to NBS was evaluated by

adjacent -range comparisons, as in [3]» with the calibration console. The

results are shown in table 12.

The 10-V range was compared with F^10-FY2, as the basis for the

build-up. However, insuring the validity of the build-up process is somewhat

more complicated than with the F^ set. Each series resistor is used with

both TE's, Fyi and Fy2, to form two ranges (such as 3 and 6V with the 800-Ji

resistor). Then, since dy-d|.+dQ, the difference between those two ranges is

^v2~^v1 "^t2~^t1 *"^c2~^c1 • Thus d^ of each of the two TE's must be evaluated

(at the comparison currents if it is current dependent). Since d^ depends on

the heater of the TE as well as the resistor, any difference between the two

d^-'s should also be evaluated to determine dv2"dvi » the correction to be

48
applied at each step of the build-up comparisons. This Is quite complicated

unless the reactance terms are negligible as shown in appendix 5B and

discussed in the next section.

Fortunately the d^ variations caused by TE's decrease with increasing

range. For this build-up of the Fj set, d^ corrections were evaluated from

the comparison of the 6-V and higher ranges at 20 Hz and 1 kHz (where the

efect of dj^ of each heater is negligible). Since the dg's of the series

resistors are also negligible at these relatively low frequencies, the same

^c2~^Q^''^y2~^v^ evaluated in each of these comparisons, where (1^2 refers

to Fyl at 1.25 or 1.67 mA, and d^,^ to Fj2 at 5 mA. The average value is less

subject to random errors. For these TVC's each (iy2~^y^ ^^^^ small. At

20 Hz and at 1 kHz the average of all of them at 6 V or above was 0 ppm.

Thus the ^q2~'^c] corrections were negligible at these frequencies. They

were taken to be the same at higher frequencies. This assumption was checked

to a few ppm by special tests involving the known F^ TVC's and the Fyl and

Fj2 TE's. Similarly, the d^ corrections caused by heater impedances were

very small.

The results of the Fj build up are given in table 13. They were checked

by comparing the 1-V and 100-V ranges with the same ranges of the F^ set.

The differences ranged from 2 ppm or less at 1 kHz to 6 ppm at 100 kHz, and

are generally within the random error limits imposed by Sp^, which were

1.2 ppm at 1 kHz and 2.8 ppm at 100 kHz.

IX. THERMOELEMENT REPLACEMENT

It is highly desirable to be able to replace a TE in a TVC set without

recalibrating all of the ranges. The analysis in appendix 5B of the

substitution error which may result shows that the ac-dc differences as current converters, d^, and as voltage converters, d^^ of both the old and new TE's must be known, and the phase angles of the impedances must be less

49
than certain specified limiting values. At NBS, d^ and dy^ can be measured

by comparison with the reference standards, and the phase angles can be

determined closely enough with a commercial impedance comparator to 100 kHz.

If the phase angles are negligible, the change in d^ of a given range of

a TVC set when TE^ is replaced by TE2 (of the same nominal range and input

resistance) is given by

^h ^s

^2"^1 " ^^h2'^h1^ * ^^C2"^c1^ •

where and Rg are the nominal values of the heater and series resistances,

Rt"Rh*^s» ^vh applies to the heater alone.

It is to be noted that as the voltage range is increased, the first term

becomes negligible and the second approaches dQ2~dci

.

If the phase angles are not negligible or cannot be determined, it is

best to repeat the lower inter-range comparisons. Although the phase-angle

term is complicated, the analysis shows that the error in neglecting it will

be less than 1 ppm if the angles of the two TE's are equal to within 0.1 mrad

and the angles of the resistors and the TE's are less than 5 mrad.

X. ERROR ANALYSIS - GENERAL REMARKS

These are personal views on various phases of error analysis.

(1) There are many sources of systematic errors in these standards and

in the comparison process. Practically all are independent. Our estimates

of their individual limits, after all known corrections are applied, should

then be added in quadrature as the square root of the sum of their squares

(rss), following Youden [10].

50
(2) Youden recommends adding the estimate of the random errors (such aa

3Sa) to the rss of (1) to obtain the total uncertainty.

(3) However, it is evidently desirable to follow the much more recent

guidelines of the BIPM Working Group on the Statement of Uncertainties, in

accordance with B. N. Taylor's memorandum of August 21, 1981 [11]. This

involves adding all of the components in quadrature, including s^. They are

to be estimated as the equivalent of "standard deviation". For convenience,

the BIPM recommendations are reproduced in appendix 6.

(4) It is generally not feasible to determine a probability distribution for each of these error estimates, even though one feels that the true

value may be nearer the center than at the estimated limits. Thus it is

safer (more conservative) to assume a rectangular distribution, with limits

±b. The standard deviation, a, of such a distribution is b//3.

C. F. Dietrich, of the British Calibration Service, has shown that if rectangular distributions are combined with each other and with a normal (Gaussian)

distribution, the standard deviation of the combination is

n -

As a result of his extensive analysis of such combinations he states that

[12]

"M.69 Let us look at this another way round. If we know

the standard deviations of the Gaussian uncertainties of an Item of calibration, and thus the total Gaussian standard

deviation, and if we assess the maximum value of each of

the estimated uncertainties, and assume that these have a rectangular distribution, we can at once find the total

standard deviation of the measurements. We can then state

that the probability of an uncertainty lying outside ±2a is

less than 0.0455 or the probability of an uncertainty lying

outside ±3o is less than 0.0027, without having to go to

all the trouble of calculating the combined distribution

and calculating the required probabilities. The correct

51
probabilities will always be less than the corresponding

Gaussian ones having the same tolerance limits and standard

deviation, but usually by only a small amount."

This very important conclusion provides a simple yet generally realistic

method of combining error estimates. It is on the conservative side (i.e.,

does not underestimate the total).

(5) A very important factor in error analysis is to gain enough

knowledge of, and experience with, the measurement process to be able to find

the (generally hidden) causes of residual uncorrected errors, and to get

reasonable estimates of the error limits. This inevitably takes time.

(6) Because of the difficulty of finding error sources by simply

repeating measurements in the same way, it is very desirable, as is well

known, to vary the conditions of the measurement and note the effects on the

results. The imposed variations should be well beyond the normal limits.

(7) Thus in research to establish the accuracy of physical standards,

every effort should be made to vary or change the associated measuring

equipment. In contrast, as Churchill Eisenhart has emphasized [13], the use

of standards in calibrating others should be considered as a measurement

process which must be kept under statistical control, with only specified

ranges of the variables allowed. The measurement process should be well

documented to insure this through the years.

(8) It is also very desirable in the research phase to make

measurements with different standards and different comparison apparatus,

whenever possible. We know that having different TE's made in different ways

by different manufacturers can give a lot of confidence in the results if

all of them agree, and can help disclose sources of error if they do not.

52
(9) "Round-Robin" teats of three or more TE's or TVC's of the same

range, and other more complex experimental designs, can also be very valuable

means of disclosing systematic errors, as is well known.

(10) As figure 1 shows, there are several steps in the determination of

the accuracy of the final or working standards of ac-dc difference. It is

well to note that, if corrections are applied at each step for the known

(determinate) errors, the estimates of remaining residual errors (these are estimates not errors) are combined as the square root of the sum of their

squares (rss), not added directly. There is no need for the crippling

"accuracy ratios" or factors of 3/1 to 10/1 in each step, as so often

specified.

(11) Operator mistakes and certain equipment malfunctions are not

amenable to analysis and inference, but can be major sources of error. For

established calibration programs, where the volume of work justifies it, they

can be reduced by computer control and computer verification of important

parameters. They can also be reduced by repeating critical calibrations with

a second operator and a second calibration console, and by the calibration of

check standards in an adequate quality- control program. Having a second

comparator and equipment available has been very helpful in the ac-dc

program, in both the research and calibration phase.

(12) Incorrect procedures, generally stemming from incomplete

understanding, can also be major sources of error. As F. B. Silsbee has

emphasized [14], the person in charge of a standardizing laboratory must have

a high degree of technical knowledge and competence. This is especially true

for ac-dc difference measurements, which are deceptively simple in principle

but require a person with a sound knowledge of ac circuit theory to detect

and overcome problems.

(13) It is to be expected that the ranges of the reference standards

closest to those of the primary standards will be known with the best

accuracy, and that the accuracy will decrease as both range and frequency are extended. Thus a realistic view of the uncertainty, u, will be that of a

53
rather flat-topped cone in a S-dimensional space, with the range and

frequency as x and y coordinates and 1/u as the z (vertical) coordinate. The

views for the working standards will, of course, be similar.

XI. RANDOM (STATISTICALLY EVALUATED) ERRORS

A. MJTC Comparator

A limiting factor in the precision of TE comparisons was the noise level

of the circuit and nanovoltmeter , With the 10-s averager for each reading

and with 120 s required for an ac-dc determination, the pass band of interest

extended from about 0.01 to 0.1 Hz. Both the noise level and the response

time of the nVM were dependent on the source resistance. With these factors

in mind, a number of tests were made to evaluate the effects of noise and

zero drift of the comparator and its nVM on ac-dc measurements, with the two

thermocouple inputs shorted.

The first of these consisted of eyeballing the peak-to-peak excursions

of the nVM for 10 s, while at the same time taking 10-s averages of the nVM

output (expressed in nanovolts of input). The average difference between

these 10-3 averages was generally 1/8th to l/20th of the average of the

peak-to-peak excursions, a marked improvement. However a better measure of

the effect of noise on the integrated output was obtained by calculating a phantom A' - ^T'ds, by the formula given in section III, using groups of

successive 10-s readings and assuming nE - 20 mV. For both the 300 and

1000-nV ranges of the nVM usually used, the average A' (without regard to

sign) was less than 0.1 ppm.

Similar (although fewer) measurements were made with maximum source

resistance (300S2), and with 30 s between averages, as in a normal ac-dc

comparison. The values of A' ranged up to 0.15 ppm. They would be expected

to be 3.3 times as great with emf's of 3 mV, as in the TVC build-ups. These

54
are the true limiting minimums of random errors imposed by the comparison

process.

B. Other Conparatora

Similar tests were not made for the 1966 comparator and for the

calibration console. Both have been in use for long periods of time.

Instead the pooled standard deviations of each were evaluated under the

conditions of the present investigation, for TE and TVC comparisons, as discussed in the next sections.

C. Statistical Parameters and Tests

Four independent determinations were normally made at each frequency in

each comparison of the TE's, and the average, A^, and its standard deviation,

Sg^, were computed. Since the same measurement process was involved in many

such tests, pooled standard deviations were computed to increase the precision of estimating the true a. If each of m comparisons has the same number

of determinations, the formula for the pooled standard deviation of the

average is simply

s .(lis 2)1/2 . pa m ^ ai

Preliminary calculations were needed to determine which tests to pool.

Obviously comparisons at low emf's were likely to have larger s^'s than those

at 10 mV. On the other hand tests at different frequencies rarely had

significantly different s^'s*

The pooled "within-day" s^'s were used to more sharply determine the

significance of differences between averages of measurements repeated on different days. For two measurements the "t" test, with the formula given in

section III, was simplest. For more measurements, appropriate tests were

used [15].

55
The round-robin triads of figures 3 and 1, the build-up comparisons for

the TVC's, and the ccmiparisons of ampere-range TE's with shunted TE's at low

frequencies were all examined for evidence of systematic errors.

D. Pooled Standard Deviations

Table ^^ shows the values of the pooled standard deviations, Sp^, for

the several paths in the Road Map of figure 1 . Each was grouped by

parameters, such as type of comparator, frequency, and emf level, that were

found to be significant.

The table shows the wide range of Sp^'s from the MJTC's to the working

standards. These values are combined in section XIII, with the systematic

error estimates of section XII.

E. Experimental Designs and Tests

The sequence of readings in a determination (ac, dc+, dc-, ac, equally

spaced in time) is an example of a simple experimental design that eliminates

the error from linear drifts in the TE's and comparator. Knight and his

coworkers at NPL have used a least-squares fit to the readings in multiple

ac-dc sequences to correct for 2nd and 3rd order drifts as well. However the

excellent temperature stability in MET A1ll4 at NBS, and the long thermal

time-constant afforded by the thermally- insulated tank in which the TE's are

placed, have made this unnecessary for this investigation. Erratic changes,

chiefly from the nVM, would generally mask small non-linearities.

Experimental designs based on left-right interchanges of two standards

to detect offsets could not be used directly, because of the often large

differences in the emf's of the TE's (as great as 3/1). With the MJTC

comparator, the higher emf must be connected to the divider, and the output

of the TE with the higher n must be connected to the potentiometer.

56
Triads of comparisons (round robins) are shown In figures 3 and 4. In

each of these there are two paths from one TE to a second; one direct and the

other via the third TE. If the d2-di paths differ by more than (2/3)Spg^,

there Is evidence of systematic error at about the 95% confidence level.

None of these 28 triads showed such evidence. Other more complicated round

robins, used to evaluate TE's after the loss of FX and FY, also showed small

closure errors.

In the 50 to 5 mA comparisons of figure 3, the values of d for FY and

FZ, obtained after 4 stages from 0F19, were within 1 ppm of those obtained

from the direct MJTC tests at 1 kHz and the subsequent frequency extensions.

XII. ESTIMATES OF RESIDUAL SYSTEMATIC ERRORS

(HOT STATISTICALLY EVALUATED)

There are many known sources of residual errors in the TE's, TVC's and

comparator. A listing of those which are significant and of the estimates of

their limits (bounds) is given in ppm (and percent of A - d^-d3 where

applicable) in table 15 for each of the blocks in the Road Map of figure 1

.

The estimates of limits are given for 1 kHz, where many errors are at a minimum, and for the highest frequency. Unless otherwise noted, equal bounds

are implied; i.e., 0.5 signifies that it is considered to be very unlikely

that the residual error will be outside the range -0.5 to +0.5 ppm.

Many of the bounds were determined from actual measurements or specifications. Others were difficult to estimate, and a few are simply

educated guesses; based, however, on extensive experience in ac-dc

measurements. Some pertinent notes are given at the end of the table.

Almost all of the sources of error are independent (not correlated). The few

correlated sources are discussed in the next section.

57
XIII. COMBINATION OF UNCERTAINTIES

From the basic dimenslonless equation given in section III-A

ci,p = dg + (N(j-N^)/ngE2^. Here the exponent of each variable is +1 or -1 . From the theory of errors [16], the propagation-of-error formula for the

fractional standard deviation, s^,, of d^, with uncorrelated variables having

small random errors, reduces to (Z s^^^)^'^^, where each of the Sj^'s is the

fractional standard deviation of an individual factor (considering (N^j-N^) as

a factor). In accordance with Dietrich [12], the same formula will apply

safely enough for combinations of estimated residual uncertainties^, each

taken to be uniformly distributed between estimated bounds, ±b, with the

standard deviation of each taken as s^^ = ±b^//3. These values of Sj^ are the

"equivalent standard deviations" of the BIPM report (appendix 6). Twice the

overall standard deviation, 2Sq, is taken as a reasonable measure of the

overall uncertainty. According to Dietrich the probability of a value lying

outside this will be less than 0.05.

This use of 2Sq for the overall uncertainty, Uq, is strongly recommended

by Dunn of the National Research Council of Canada [17], who also recommends

the approach taken by Dietrich to the combination of uncertainties. It seems more appropriate than 3Sq when many of the sources of error are considered to

be of uniform probability density, since the long tails of the normal

distribution are truncated.

Because of the several successive steps and the three different

comparators used in evaluating the NBS reference and working standards, and

because of the many steps in the extensions of the frequency and other

ranges, the rather straightforward procedure of combining uncertainties from

tables 14 and 15 becomes very involved. The detailed calculations are shown

in table 16. They are generally self-explanatory. The results are the same

as those obtained by adding variances and finding the overall uncertainty as twice the square root of the overall variance. However, no definite

confidence limit is associated with the uncertainty, U.

with each other and with s„. See section X-i|.

58
Extreme rigor in making these calculations did not seem warranted,

especially for the smaller U's, which do not effect the rss results very

much.

In the tvC build-up the errors in determining the changes in the

of the TE's (FX2 and FY2) with current level are correlated; i.e., they add

in each step of the build-up. Similarly in the Fy TVC build-up, the errors

in determining the difference between the d^'s of Fy! and ¥^2 at the

particular currents used add at each step. These correlated uncertainties

were separately evaluated in the calculations of table 16.

It is to be noted that in each build-up involving chains of successive

measurements the uncertainty at the end of each chain is given in table 16.

It is difficult to estimate the effects of the uncertainties which

depend on the magnitude of A, particularly for these chains of measurements

These range from 2.8$ of A for the MJTC comparator to 3.6/6 for the other

comparators. A prudent allowance for them is given in B of table 17.

Fortunately the values of d for the MBS standards are small, so that this i; a significant factor in only a few cases.

The values of d and U apply at the currents and voltages shown in the

various tables. At other voltages and currents additional corrections must

be applied to the d*s, and the uncertainties will be somewhat greater.

Appropriate currents and voltages were used in evaluating the reference

standards. Fortunately the working standards are rarely used at other than

rated inputs.

59
XIV. CONCLUSIONS

A. Uncertainties of the NBS Standards

The overall results of the calculations of the uncertainties are given

in the important summary table 17. They apply at rated current and voltage,

with the exceptions shown.

Several conclusions can be drawn from this table and the other results

of this study, as follows:

(1) The long-term stability of the d's of the MJTC's is excellent.

This, together with their low d's and 0.5 ppm uncertainty, amply fulfilled

the hope that they would be superior primary standards of ac-dc difference at

the base frequency of 1 kHz, and over the range from 30 Hz to 10 kHz.

(2) Their accuracy can be transferred to the single-junction TE's at

5 mA and 5 to 10 V to 1 ppm.

(3) The accuracy of these base ranges can be extended to 100 kHz to

better than 2 ppm.

(U) As expected, (see section X-13) the uncertainty of the reference

standards increases with increasing V or I range. The most precise

comparator can only be used to 50 mA or 100 V, so the uncertainty cone has

discontinuities in slope.

(5) As expected, the difficulties of extending the range increase at

higher frequencies, and so do the uncertainties; generally by a factor of 1 .5

to 2 times those at 1 kHz.

(6) As shown in table 16, the random-error component is significant in

almost all comparisons. There is generally a reasonable balance between

60
random and systematic components. However, better precision would be the

first step to improving accuracies.

B. Calibration Uncertainties

(1) If TE's and TVC's of the same type and quality as the working

standards are calibrated with the calibration console, it would be expected

that the overall uncertainty would be the rss of the working standards from

table 17 and the single step U's of 16-H (minus the Sg of the reference

standard) or 16-1 of table 16. The results of such calculations are given in

table 18, along with the present calibration allowances, from section I-C.

(2) As the table shows, little improvement may be possible for the

coaxial TVC calibrations. For TCC calibrations the differences between the rss uncertainties and the allowances are deceptive. Individual TE's of the same quality as our standards are no longer available in the ampere ranges.

The multirange TVC's with shunts that are comraercially available are much

more complex. Experience has indicated that they have larger and less stable

d's than do the single-range TE's and coaxial TVC's.

(3) If NBS is pressed for better accuracy, particularly for TVC's, the

needs might be met by using the uncertainty-cone concept to specify smaller

uncertainties at and near the apex. In this way the basic accuracy at NBS

could be transported via suitable ac-dc transfer standards to the client's

standards laboratory.

Very high accuracies are attainable over limited ranges for very

special tests (such as international comparisons) In which such accuracies

are really needed. They would be made by directly comparing single-range

coaxial TVC's up to 100 V with the resistors and reserve TE's (FA and FB),

or single-junction TE's up to 50 mA with one of the pairs of reference TE's,

using the MJTC comparator in each case. The results of calculations (similar

to those described in B-1 of this section) are shown in table 19, to

substantiate this.

61
(5) A comparison was made of the d's of the working standards shown in

tables 11 and 13 with the values previously assigned from earlier tests and

presently used in calibration work. For the TVC's, the largest difference

between the d's was 23 ppm, on the 600-V range at 100 kHz (the highest

voltage available at this frequency). It was 10 ppm on the 1000-V range at

50 kHz, and much less at most other ranges and frequencies. The average

difference for all ranges at all frequencies (with regard to sign) was only

2 ppm. For the current-measuring TE's the largest difference between the d's was 36 ppm. The average for all ranges at all frequencies was only 8 ppm.

These low differences attest to both the quality of the earlier measurements

and the stability of the NBS ac-dc transfer standards. They are comfortably

below the uncertainties of calibrations shown in section I-C of this report.

XV. RECOMMENDATIONS

A. Maintenance of AC-DC Standards

(1) The NBS standards are believed, with reason, to be very stable.

However if better accuracy than heretofore is to be maintained, it would be

prudent to compare the working standard TE's with the reference standards

every three years, and to compare adjacent ranges of the Fj TVC set at

similar intervals. This could be modified by experience — see recommendation B-6.

(2) The difficult and very time-consuming recalibrations of the

reference standards might only be necessary every 9 years, if the 3~year

tests show no significant changes.

(3) Spare Fyl and Fj2 modules should be available in case of burnout,

with values d^ and d^ known by prior comparison with the present ones. FA

and FB, an extra pair of 5-mA TE's with known d^ and d^, are already

available for the F, reference set. The substitution formulas of section IX

62
should make it possible to calculate the d^'s for the other TVC ranges,

without recalibrating each one, when a TE is replaced.

(M) AC-DC differences as large as 9 ppm have been observed even in

Evanohm-heater TE*s, so replacements should never be used without

recalibration.

B. Quality Control of the Calibration Process

(1) The principles of quality control given by Eisenhart [13] should be

followed. Check standards should be evaluated periodically in the

calibration console. They should be of the same types as the major kinds of

ac-dc standards normally calibrated. In particular the multirange standard,

which was purchased in part for this purpose, should be put in service, as well as the spare Model F TVC's on which a partial history has already been

garnered.

(2) NBS statisticans should be consulted and should be of major help in

setting up these programs. Careful selection of a few ranges and frequencies

should not make these very important quality-control tests too time-consuming

or onerous.

(3) The waveform and frequency of the applied ac should be observed on

a calibrated CRO in each test, and the frequency setting of the oscillator

should be double-checked to avoid mistakes.

(M) The very simple EMI test described in section III-A should be made

occasionally, especially for low ranges.

(5) A second comparator and associated equipment should always be

available and ready for use as a check against unsuspected (or suspected)

errors. This is particularly valuable when new or modified ac-dc standards

are first tested.

63
(6) As experience with the check standards and the bienniel

calibrations of the working standards is accumulated, the recommended

calibration periods of A-1 for the working and reference standards may safety

be revised.

C. Other Recommendations

(1) Almost all of the TE's from 5 mA to 20 A were made to our special

order by American manufacturers who no longer make TE's. The MJTC's are not

commercially available at all. All of these now form an unique collection of

well-evaluated reference standards, and should be reserved solely for

evaluating the NBS working standards and for comparisons with other national

laboratories, or for other special test in which the highest accuracy is

really required. The MJTC comparator, with its high precision, should be

available for use when this is really needed.

(2) The new types of solid-state thermoelements (SSTE's) have much

higher emf's than the conventional TE's, but are subject to the same basic

limitations on ac-dc difference from Thomson, Peltier and other effects. If

Wilkins' -type MJTC's can not be obtained commercially in the near future,

serious consideration should be given to designing and making thin-film

MJTC's by cooperation with other divisions at NBS.

(3) The new values of d for the working standards, from tables 11 and

13, should be used in place of those now in the computer. However any stated

improvement in the calibration uncertainties should be withheld until some checks on the error analyses are made by comparing the NBS working and

reference standard TE's and TVC's In both the calibration and 1966 consoles.

In addition any improvement in the stated uncertainty of the more complex

multirange TVC's and their shunts should await the installation of a quality-control program and further studies of these instruments.

64
XVI. ACKNOHLEDGEHENTS

Sincere thanks are expressed to J. R. Hastings, E. S. Williams,

C. B. Chllders, and K. B. Krlshnaourthy, who assisted In this Investigation,

to N. Beleckl for his encouragement and support, to C. Reeve for his helpful

suggestions and review of the statistical procedures, and to T. Llpe for his

review of the entire report.

65
REFERENCES

1 . Thermal Converters as AC-DC Transfer Standards for Current & Voltage

Measurements at Audio Frequencies, F. L. Hermach, Jour. Res. NBS, v. 28,

no. 2, p. 121 , Feb. 1952.

2. Thermal Converters for Audio-Frequency Voltage Measurements of High

Accuracy, F. L. Hermach and E. S. Williams, IEEE Trans, on Instrum. &

Measmt., v. IM-15, p. 260, Dec. 1966.

3. Thermal Voltage Converters & Comparator for Very Accurate AC Voltage

Measurements, E. S. Williams, Jour. Res. NBS, v. 75C, p. 1M5, July-Dec.

1971 .

H. An Investigation of Multijunction Thermal Converters, F. L. Hermach and

D. R. Flach, IEEE Trans, on Instrum. & Measmt., v. IM-25, p. 524, Dec.

1976.

5. Thermal Voltage Converters for Accurate Voltage Measurements to 30

Megacycles per Second, F. L. Hermach and E. S. Williams, AIEE Trans., v. 79-1, p. 200, July I960.

6. Thermal Current Converters for Accurate AC Current Measurement, E. S.

Williams, IEEE Trans, on Instrum. & Measmt., v. IM-25, p. 519, Dec.

1976.

7. Errors in AC-DC Transfer Arising from a DC Reversal Difference, B. D.

Inglis, Metrologia, v. 17, p. Ill, 1981.

8. Theoretical Analysis of the AC/DC Transfer Difference of the NPL

Multijunction Thermal Converter Over the Frequency Range DC to 100 kHz,

F. J. Wilkins, IEEE Trans, on Instrum. & Measmt., v. IM-21 , p. 33^, Nov.

1972.

66
9. Evaluation of AC-DC Transfer Errors for Thermal Converter-Multiplier

Combinations, B. D. Inglls, Metrologla, v. 16, p. 177, 1980.

10. Uncertainties In Calibrations, W. Youden, IRE Trans. Instrum. , v. 1-11,

p. 133. Dec. 1962.

11. Uncertainty Assignment, B. Taylor, Memo for Technical Staff of the NBS

Electricity Division, Aug. 21, 1981.

12. Uncertainty, Calibration, and Probability, C. F. Dietrich, Halsted

Press, 1973.

13. Realistic Evaluation of the Precision and Accuracy of Instrument

Calibration Systems, C. Elsenhart, J. Res. NBS, v. 67C, p. I6I,

Apr-June 1963.

1M. Suggested Practices for Electrical Standardizing Laboratories, F. B.

Sllsbee, NBS Circular 578, 1956.

15. Experimental Statistics, M. G. Natrella, NBS Handbook 91, 1966 (Chapters

3 and 15).

16. Notes on the Use of Propagation of Error Formulas, H. H. Ku, J. Res.

NBS, V. 70c, p. 263, Oct-Dec 1966.

17. Measurement Assurance, A. F. Dunn, NRC, Canada, Report 188^43, 1980.

18. A Resistor Calculable from AC to w=1o5rad/s, R. J. Haddad, Thesis, GWU

School of Engineering and Applied Science, April 1969.

19. The Theory of Peltier & Thomson Errors In Thermal AC-DC Transfer

Devices, F. C. Wlddls, lEE Monograph M97M, Jan 1962.

I

67
Table 1

Coaparlsons of Four HJTC*s

S 30 Hz 1 kHz 10 kHz Date

G44 G5C +0.47 +0.15 -0.M4 12/08/81

(B10-41I) (B5-2)

-0.10 12/11/81

-0.02 12/11/81

Gr25-1 G5C +0.32 0.00 -0.29 12/11/81

(C10-1) (B5-2)

-0.03 12/U/81

Gr25-1 G4l» +0.17 -0.01 -0.05 12/14/81

(C10-1) (310-44)

0.06 12/18/81

NPL 14 G44 +0.34 12/18/81

(A50-14) (310-44)

+0.36 12/18/81

-0.40 +0.27 12/18/81

Gr25-1 NPL 14 +0.52 -0.22 12/18/81

(C10-1) (A50-14)

68
Table 2

Differences Between 1981 and 1976 Evaluations of MJTC*8

**81 - ^76 (PP")

MJTC

044

Gr25-1

G5C

G10C

NPL 14

30 Hz

+0.2

0.0

+0.3

-0.3

-0.2

1 kHz

0.0

0.0

-0.1

0.0

0.0

10 kHz

+0.1

-0.4

+0.1

0.0

69
Table 3

Results of Comparisons of FX and FY with NJTC*s

TE

or

TVC

FX

FX

Fi5-FX^

Fl5-FX

FY

FY

F^IO-FY

F^10-FY

IorV

5 mA

2.5 mA

5 V

3 V

5 mA

3 mA

10 V

5 V

30 Hz

+0.9

-0.1

+0.8

+0.2

d(ppn)

1 kHz

-0.3

0.0

-O.M

-0.7

+0.2

+0.4

-0.3

+0.1

10 kHz

-0.2

-0.1

-1 .2

-1.0

0.0

-0.2

-0.8

-1.2

^F^5-FX consists of the 5-V resistor of the F^TVC set, in series with TE FX,

70
Table 4

Results of TVC Tests with Twin Resistors

<lT-ds (ppm)

T S 30 Hz 1 kHz 20 kHz 50 kHz

FY FX +0.2 +0.2 0.0 -0.2

B5E77 FX +5.0 +2.6 +3.2 +3.7

F5K9 FX -1 .6 -1 .0 -1 .1 -1 .2

B5EXG1 FX +2.8 +1.1 +0.9 + 1.5

A5EP7 FX -0.1 +0.5 -0.5 -1.3

Average d of 6 TE's

with - 0.0 +0.6 +0.4 +0.4

NOTE: All tests at 5 mA.

30 Hz results not averaged. See text.

,1

j

71
Table 5

Anpere-Range CoBparlsons

V^s

I

T S (A) 20 Hz 1 kHz 20 kHz 50 kHz

W20A#7-1 W10A #0-1 10 ••1

1

17 +10

W10A#0-1 W5A #5-2 5 -5 -2 0 +16

W10A#0-1 W5A #5-1 5 -6 -6 -6 +17

W5A #5-2 W3A #4-1 •fi -1 4 +16

W3A W2A #3-2 2 *3 0 -7 -11

W2A #3-2 W1A #7-2 1 -4 -2 -5 -13

FG3A W3A #4-1 3 -2 0 15 +43

FG3A W3A #4-1 2 -4 1 15 +46

FG1A W1A #7-2 1 -3 -2 11 +29

FGIA W1A #7-2 0.6 +1 0 15 +36

FH2A W2A #3-2 1.2 -2 -2 -12 -40

FH2A W2A #3-2 2.0 0 -14 -33

FH20A W20A #7-1 10 -21 -49

FH20A W20A #7-1 16 -22 -59

NOTE: FGIA, FH2A. FG3A, and FH20A are Williams' shunts with low-range TE'

FG and FH.

72
Table 6

AC-DC Differences of 1A to 20A TE*s

d(ppin)

I

TE (A) 20 Hz 1 kHz 20 kHz 50 kH

W20A #7-1 10 -5 +5 +14 +10

W10A #0-1 5 -6 -6 -3 0

W5A #5-2 3 -] -^ -3 -16

W3A #4-1 2 -2-3 -7 -32

W2A #3-2 1 -5-3 0-21

W1A #7-2 1 -1 -1 +5 -8

W1A #2-2 1 -4-3 +2 -10

73
Table 7

Results of Evaluation of FX2 and FY2

TE I d(ppa) or TVC

orV 20 Hz 1 kH^ 10 icHz 20 kHz 50 kHz 100 kHz

"1 ov +3.9 +2.H +1.3 +1.0 +0.6 +0.3

7 +2.0 + 1 A +0.6

' +1.6 tl.il 0.0

FY2 5 mA +4.7 +3.2 +3.6 +3.5 +3.9

J* J +? ft + ? Q

2.5 +2.2 +2.3

5 mA +3.5 +2.2 +2.M +2.1 +2.2

3^3 • +2.3 + 1.3 + 1.3

2v5.- , +1.6 +1.9

74
Table 8

Conparlaona of F^tvc*8

(FXg and FY2 TE'a)

A - dy-ds (ppn)

T S V 20 Hz 1 kHz 20 kHz 50 kHz 100 kH:

F2OO-FX2 F1 OO-FY2 100 - -1.1 +0.2 +1.4 +17.8

FIOO-FY2 F5O-FX2 50 - +0.6 -0.8 -2.8 -6.2

F5O-FX2 F3O-FY2 30 - -2.5 -1.6 -2.0 -1.5

F3O-FY2 F2O-FX2 20 -1.3 +0.7 -0.3 0.0 -2.0

F2O-FX2 F1 O-FY2 10 -3.8 -1.7 -2.1 -2.9 -6.0

FIO-FY2 F5-FX2 5 -0.6 +0.2 -0.3 -1.3 -2.5

F5-FX2 F3-FY2 3 -1.6 -0.6 -0.8 -3.1 -6.3

F3-FY2 F2-FX2 2 0.0 +1.4 +0.7 -1.1 -4.1

F2-FX2 F1 -FY2 1 +0.4 +1.9 +0.7 -3.9 -13.1

FI-FY2 FX2 0.5 +1.0 +3.4 +4.2 -0.7 -6.1

FXg FY2 0.5 +1 .2 +1.1 +1.9 +2.2 +2.5

Coipariaons with F^i

,

FX, and FY in place of FX2 and FY2

FIOOO-F7I F500-FX 500 -2.0 -15.1 -1.7

F500-FX F300-FY 300 -0.8 -0.3 -4.1 -33

F300-FY F200-FX i200 -0.7 -0.5 -2.3 +6.7

75
TVC

Table 9

AC-DC Differences of TVC 'a with FX2 and FY2

d(ppm)

V 20 Hz 1 kHz 20 kHz 50 kHz 100 kHz

+2 7 -6 1

500 +2.5 +1.2 -6.5 -18.6

300 + 1 .7 +0.4 -7.3 -19.4

300 +3.5 +1.9 -1.8 +15.6

200 +2.7 +1 .

1

-2.6 +1 4.8

200 +2.4 +0.4 -1.7 +6.4

100 + 1 .6 -0.4 -2.5 +5.6

100 +2.7 -0.6 -3.9 -12.2

50 +1 .9 -1 .4 -4.7

50 +1.3 -0.6 -1.9 -6.8

30 +0.5 -1 .4 -2.7 -7.6

30 +2.6 +3.0 +0.2 -0.7 -6.1

20 +0.7 +2.2 -0.6 -1 .5 -6.

9

20 +2.0 +1.5 -0.3 -1.5 -4.9

10 +0.1 +0.7 -1 .

1

-5 7

10 +3.9 +2.4 +1 .0 +0.6 +0.3

J + 1 H +1 6 +0.2 -0 2 -0V.J5

5 +2.0 +1.4 +0.5 +1.1 +2.0

+0.7 +0;6 V.J +0.^ V.J + 1 .2

+2. +1 .2 +0.5 +'^ . 4 +7 .

5

2 +0.4 +0.4 -0.3 +2.6 +6.7

2 +0.4 -1 .0 -1.0 +3.7 +10.8

1 -1 .5 -1.8 -1.3 +3.9 + 10.0

1 -1.9 -3.7 -2.5 +6.8 +23.1

0.5 -4.4 -4.5 -3.3 +6.0 +22.3

0.6 -5.4 -7.9 -7.5 +6.7 +28.4

0.6 -6.6 -8.9 -9.4 +4.5 +25.9

F1000E-Fyl

F5OO-FX2

F3OO-FY2

F2OO-FX2

FIOO-FY2

F5O-FX2

F3O-FY2

F2O-FX2

FIO-FY2

F5-FX2

F3-FY2

F2-FX2

F1 -FY2

FXg

FYg

76
Table 10

Coaparlsons of Working and Reference Standard TE*3

A(ppm)

I

Test Standard (mA) 20 Hz 1 kHz 20 kHz 50 kHz

A3 #95 FA 5 +5 +2 +9 +18

G10 #57 B10EP2 10 -5 "5 "4 -3

F20 #1 W25E1 20 -9 "7 "5 +1

W30 #1 W25E1 25 -12 -11 -12 -12

F50 #2 W50E1 50 -37 0 +3 "2

B100 #3 W100E1 100 -18 -20 -20 -2M

A250 #4 W250E1 250 -19 -22 -28 -44

(A)

W0.5A #0-2 W0.5A #0-1 0.5 -4 -2 -5 -2

W1A #7-1 W1A #2-2 1 0-2-1 -4

W2A #8-3 W2A #3-2 2 +3 +1 +1 "2

W3A #9-2 W3A #4-1 3 +2 -1 -1 S

W5A #5-1 W5A #5-2 5 -5 -7 -11 -13

25 Hz

W10A #0-3 W10A #0-1 10 -6 -5 -13 "28

100 Hz

W20A #1-2 W20A #7-1 16 +3 "10 -18 -14

77
TE

Table 11

AC-DC Differences of Working Standard TE*3

AC-DC Difference (ppm)

I

(mA) 20 Hz 1 kHz 20 kHz 50 kHz

A5 #95

G10 #57

F20 #1

W30 #1

F50 #2

BlOO #3

A250 #1|

W0.5A #0-2

W1A #7-1

W2A #8-3

W3A #9-2

W5A #5-1

W10A #0-3

5

10

20

30

50

^00

250

(A)

0.5

1

2

3

5

10

2

-6

-10

-13

-36

-20

-2H

0

-i»

-2

0

-6

25 Hz

-12

100 Hz

0

-6

-8

•12

-1

•21

•22

-2

-5

-2

-4

-11

-11

+8

-5

-6

•13

+2

20

28

-3

+ 1

+ 1

-8

-U

-16

+17

-4

0

-13

-3

-23

-43

-7

-14

-23

-35

-29

-28

W20A #1-2 16 +8 -5 -4 -4

78
Table 12

CoBparisona of Fj tvc*s

A - dj-ds (pp«)

T

e

V £U HZ 1 Knz £U Knz 7U Knz 1 UU KnZ

Fyl 000-1 OE-Fyl « ^AA n A Fy600-FY2 600 0 -13 -7

Fy500-10C-Fy1 F7600-F72 500 0 -7 +1

1

FySOO-IOD-Fyl Fy600-Fy2 500 0 0 +35

FySOO-Fyl F7200-F72 200 -1 -1 0 + 12 +59

FylOO-Fyl F760-F72 60 -1 +1 0 -6 -13

FySO-Fyl F720-F72 20 +1 0 0 +1 -1

FylO-Fyl F76-F72 6 +2 0 -1 +3 -2

F73-F71 F72 2 +2 -2 -1 -1 -3

F71 F72 1 1 -1» -1 -3

79
Table 13

AC-DC Differences of Fy TVC 8

d(pp«)

TVC V 20 Hz 1 kHz 20 kHz 50 kHz 100 kHz

Fyl 000-1 0E-Fy1 600 - -2 -15 -2 -

FySOO-IOC-Fyl 500 - -2 -9 16 -

Fy500-10D-FYl 500 - -2 -2 il0 -

Fy600-Fy2 500 -2 -2 +5 +43

F73OO-F7I < 300 -u -2 -2 +5 +43

F72OO-F72 200 -3 -1 -2 -7 -16

FylOO-Fyl 100 -3 -1 -2 -7 -16

Fy60-F72 60 -2 -2 -2 -1 -3

FySO-Fyl • 30 -2 -2 -2 -1 -3

F720-Fy2 20 -3 -2 -2 -2 -2

F^IO-Fyl 10 -3 -2 -2 -2 -2

Fy6-Fy2 6 -5 -2 -1 -5 0

F73-F7I -5 -2 -1 -5 0

2 -7 0 0 -4 +3

F7I 1: -6 -H -4 -5 0

80
Table 14

Pooled Standard Deviatlona, Spa

_Spg (ppa of input V or I)

Type of Teat

Typical

(aV) 20 Hz 1 kHz 20 kHz 50 kHz 100 kHz Average

1. Comparison of 10-^40 0.11 0.16

MJTC'3 (5)^ (11)

Comparison of

Reference TE's

and F1IOFY2

10 0.20 0.29

Extension of

of frequency

range

10 0.23

(5)

0.211

(5)

0.25

(5)

DUXXU up

Reference TE's

a. 5 to 50 mA 10 0.26 0.23 0.35 0.26

fill

-

b. 50 mA to 1A 10 1.80

(11)

1.68

(16)

1.57

(15)

1.98

(17)

-

0 1 A to 20A c 1'•II77

(13)

1 Rfi (12)

1 U8

(12)

11 . 71* 1 ^ (16)

Build upF^TVC's

a. 0.5 to 100 V 3

0.i»9

(7)

0.117

(12)

0.52

(12)

0.57

(10)

0.54

(10)

b. 200 to 1000 V 3 1.71

(14)

i.m

(H)

1.06

(3)

1.55

(2)

Comparison of

Reference and

Working TE's

5 mA - 20 A

10 2.36

(12)

2.77

(12)

2.47

(11)

1 .72

(17)

Build up

Fy TVC'3

1 to 1000 V 3 1.58

(6)

1 .20

(9)

2.16

(9)

2.511

(9)

2.79

(9)

0. u

0.25

0.24

0.28

1.76

1 .64

0.52

1 .43

2.33

2.05

1 Values in parentheses are the number of tests.

81
Table 15

Estimated Limits of Residual Systeaatic Uncertainties

(not evaluated by statistical means)

(See notes at end of table)

Limits^ ±b

1 kHz 100 kHz

ppn S of A ppm % ot t

A. MJTC Comparator & Equipment

1 . nVm & averager (2$ of - 2 - 2

indication)

2. Determination of ng - 2 - 2

3. n-Corapensation 0.1 0.1

M. DC-reversal difference 0.1 0.1

definitions

5. AC effects 0.1 0.2

\ - (Lb^)^''^ 0.17 2.8 0.25 2.8

B. 1966 Comparator & Equipment

1. 1966 evaluation 1.0 - 1.0

2. Detector accuracy - 3

~ 3

3. Determination of ng - 2 - 2

l». AC effects _j_ 2_^

bo 1.0 3.6 2.2 3.6

C. Calibration-Console

1 . Basic accuracy

2. Detector accuracy

3. Determination of ng

AC effects

bo 1.0 3.6 2.2 3.6

Limits, ±b, ppa

1 kHz 50 kHz 100 kHz

D. MJTC's

1. 1976 assignment of group

average 0.3

2. Possible changes 1976-1982 0.1

3. Assignment of G-44 0.1

bo 0.33

1.0 1.0

2.0

82
Table 15 (Continued)

Limits, ±b, ppa

1 kHz 50 kHz 100 kHz

E. 5 mA Reference TE's

1 . Residual thermoelectric

effects 0.2 0.2

2. High frequency evaluation 0.0 0.5

l>o 0.20 0.54

F. F^10Y2, Base TVC

1 . 1 kHz base 0.2 0.2

2. High frequency extension 0.0 1 .0

bo 0.2 1.0

G. Reference TE's, build-up

to 50 mA (3 stages)

1. Bead error (each stage) 0.2 0.3

2. Paralleling error (each stage) 0.1 0.3

each stage 0.22 0.42

H. Reference TE's, build-up from

50 mA to 1A (M stages)

1. Bead error (each stage) 0.2 1.0

2. Paralleling error (each stage) 0.3 1.0

3. AC effects (each stage) 0.0 1 .0

each stage b^ 0.36 1.73

I. Reference TE's, build-up

1A - 20A (5 steps)

1

.

Dependence of d^ on I (each step) 2.0 2.0

2. AC effects (induced emf's,

etc. ) (each step) 1.0 3.0

3. Proximity effect (each step) 0.0 2.0

each step b^ 2.2 4.1

J. Extension of Voltage Range

10 to 0.5 V, 10 to 200 V

ranges (5 steps each)

1 . Residual dependence of d^ on I (correlated) 0.2 0.5

each step b^ 0.2 0.5

83
Table 15 (Continued)

LlMlts, ±b, ppa

1 kHz 50 kHz 100 kHz

K. Extension of V range from 200

to 1000 V range (at 500 V)

(3 steps)

1 . Dependence of dy on V

(Avg. for 3 steps)

2.

3.

Dependence of d^ on I Transfer FX, FY^ to FXg,

^^2

each step

0.5

0.2

0.5

0.73

2.0

0.5

2.0

2.9

L. Working standard TE's 5 mA to

20 A range (by comparison with

reference stds, same range)

1 . Bead resistance and other

ac effects

2. Dependence of d^ on I

each step b^

3.0 5.0 5 mA

6.0 10.0 20 A

2.0 2.0

3.5 5.4 5 mA

6.3 10.1 20 A

M. Working Standards TVC's (Fy set)

Build-up 10V to 1000V range

(4 steps) and build-down 10V

to IV range (3 steps)

1. cIq of each TE (correlated) 1.0 3.0

2. Dependence of dp on V

(high ranges), build-up 0.0 to 0.0 to

0.5 2.0

3. Adp error (low ranges),

build-down 0.0 to 0.0 to

0.5 2.0

average, each step b^ 1.1 3.3

84
Sections

Table 15 (Continued)

NOTES:

A, B, & C % of L errors depend on magnitude of A - d-j-ds- n- determined once, but error affects every ac-dc determination.

AC effects are estimated residuals or allowances for rfi,

Induced voltages, etc.

E-1 & F-1 Main effects evaluated by comparison with MJTC. These are residuals.

G-1 & H-1 Estimate of error from finite bead resistance.

G-2 & H-2 Estimate of AG error, of paralleled TE's, etc.

I-l Variation of d^, with I was not applied in build-up,

1-3 Proximity of return lead on current distribution in TE heater.

J-1 The error In measuring Ad^ adds at each step.

K-1 Self-heating effect on d^..

K-3 Residual substitution error.

M-2 See K-1 note.

M-3 Ae of Fj TE'8. See section VIII-B.

85
Table 16

Calculation of Uncertainties

Se - Spa or b//3, - (ESe^)^''^^ "o - 2

S or U (ppn)

1 kHz 50 kHz 100 kHz

A. MJTC 044

from table 14-1 Sg-Sp^ 0.16

Sg for comparator,

table 15-A Sg-b//3 0.09

Sg MJTC's (15-D) Sg-b//3 0.19

0.53

B. Reference 5 mA TE's & F^10y2

TVC to 100 kHz

Spa (14-2 & 3)

Sg comparator (15-A)

Sg TE's (15-E or F)

0.29

0.09

0.12

0.29

0.14

0.31

0.29

0.14

0.58

"o - 2So 0.65 0.89 1.33

0.84 1.04 1.42

C. Reference TE's 10 to 50 mA

(3 stages)

Spa (l^'^si) P®"^ comparison

Sg corap. (15-A) per comparison

Sg TE's (15-G) per comparison

0.23

0.09

0.13

0.26

0.14

0.24

Because of triads, U^-O.S z 2Sq 0.45 0.61 per stage

After 3 stages Uqj-ZIUq 0.78 1.05

After 3 stages Uq-(Uo3^ * "b^)^^^ 1.1 1.5

D. Reference TE's 100 mA to 1A

(4 stages)

Spa (l4-4b) per comparison

Sg comp. (15-B) per comparison

Sg TE's (15-H) per comparison

1 .7

0.6

0.2

2.0

1.3

1 .0

After 1 stage - 0.8 x 28^ 2.9 4.1

After i| stages Uq|,-AUq 5.8 8.2

"d - * "c^)^/2 5.9 8.3

86
Table 16 (Continued)

S or U (ppm)

1 kHz 50 kHz 100 kHz

E. Reference TE's 2A to 20A (at 10A)

(5 steps)

Spa (14-4C) per step 1.6 1.7

Sg comp. (15-B) per step 0.6 1.3

Sg TE»s (15-1) per step 1.3 2.U

After 1 step Uq - zSq 4.3 6.4

After 5 steps Uqj - /sUq 9.6 14

"e - ("05^ * V)^^^ 11 16

F. F^TVC's 10 to 0.5V and

10 to 100V (6 and 4 steps)

Spa (l4-5a) each step a 0.47 0.54

Sg comp. (15-A) each step b 0.09 0.14

Sg TVC's (15-J)

(correlated) each step c 0.11 0.29

After 1 step - 23^ 0.98 1.26

After 5 steps UQ5-2((5c)2+5(a+b)2)1/2 2.7 4.2

Up - (Uqs^ + Ub2)1/2 2.8 4.4

G. F-,TVC's 200 to 1000V range

(at 500V) (3 steps)

Spa (l^-5b) each step 1.7 1.6

Sg comp. (15-B) each step 0.6 1.3

Sg TVC's (15-K) each step OJ*, 1 .7

After 1 step Uq - 2So 3.7 5.3

After 3 steps - /^U^ 6.4 9.2

Ug-CUqo^ * Up^)^''^ 7.0 10

87
Table 16 (Continued)

S or U (pp«)

1 kHz 50 kHz 100 kHz

H. Working TE's 5mA to 20A range

S (14-6) 2.8 1.7

Sg comp. (15-C) 0.6 1.2

Sp TE»3 (15-L) 5mA 2.1 3.1

20A 3.6 5.8

Reference TE's (16-B) 5mA 0.4 0.5

(16-E) 20A 5j^ 8L2

Uh - 2So 5mA 7.1 7.5

20A 15 21

I. Fr, Working Standard TVC's

10 to 1000V range (5 steps);

10 to IV range (3 steps)

^Da (''^"7) ®ach step a 1.2 2.8

Sg comp. (15-C) each step b 0.6 1.2

Sg TVC's (15-M)

(correlated) each step c 0^ 1 .9

After 1 step - 2Sq 2.9 7.2

After 3 steps Uo3-2((3c)2+3(a+b)2)l/2 18

(at IV) Uj - (Uo3^ + %^)^^^ 7.2 18

After 5 steps UQ5-2((5c)2+5(a+b)2)1/2 ,q 26

(at 500V) Uj - (Uo5^ "b^)^^^ 10 26

88
Table 17

Summary of Uncertainties of NBS AC-DC Transfer Standards

I. Uncertainties Independent of Magnitude of d

Uj^ ppm

Standard 1 kHz 50 kHz 100 kHz

MJTC (one of group) 0.53

B Reference base, 5 mA TE's & 10V TVC 0.84 1 .0 1 . 4 C Reference TE: 50 mA range 1 .1 1 .5

D 1A range 5.9 8.3

E 20A range 1

1

16

F Reference TVC: 0.5 & 100V ranges 2.8 4. 4 G 1000V range (at 500V) 7.0 10

H Working TE's: 5 mA range 7.1 7.5

20A range 15 21

I Working TVC's: 10V range 2.9 7. 2

IV range 7.2 18

500V range 10 26

II. Uncertainties dependent on magnitude of d:

For all standards Ug - 0.05d should be added to U^.

Ug < 1 ppm with few exceptions.

III. Uncertainties at 20 Hz are somewhat, but not significantly, greater

than at 1 kHz.

IV. Uncertainties at less than rated V or I require further study. However,

the working standards are almost always used at or near rated V or I

.

V. Self-heating effects at 1000V were evaluated in [3].

^Letters refer to table 16.

89
Table 18

Coiparison of Estimated Calibration Uncertainties with

Present Allowances

U (ppn)

1 IcHz

5 bA 20 A

50 kHz

5 OA 20 A

Working standard TE's (16-H)

Comparison (16-H without Sg of

ref. stds.)

U (rss)

Present allowance 5 mA - 5A

Present allowance 5A - 20A

7.1

7.1

10

50

15.0

9.2

18

100

7.5 21

7.5 J2

11 2H

100

200

IV

1 kHz

10V 500V IV

100 kHz

10V 500V

Working standard TVC's (16-1)

Comparison (16-1, one step)

U (rss)

Present allowance (for coaxial

TVC's)

Present allowance (for multirange TVC's)

7.2

2.9

7.8

20

50

3.0

2.9

4.2

20

50

10

2.9

10

20

50

18

7.2

19

50

100

7.2

7.2

10

50

100

26

7.2

27

50

100

NOTE ; This table applies to the evaluation in the NBS Calibration Console of

single-junction TVC's and single-range TE's of the same quality as the

NBS standards.

90
Table 19

Bstlmated Uncertainties for Speolal Tests

(Calibration of a TE or TVC against a reference standard,

with MJTC ooBparator)

U (pp.)

11 kHz kHz

5 mk 50 uk 5 mk 50 bA

Reference standard TE 5-50 raA 0.9 1.1 1.0 1.5

Comparison (16B & 16C without Sq

of TE's) 0.6 0.6 0.6 Q.6

rss 1.1 1 .2 1.2 1 .6

1 kHz 100 kHz

JV 10Y 100Y IV 10V 100V

F^TVC 0.5 - 100V 2.8 0.9 2.8 11.4 1.4 4.4

Comparison (16-C, all ranges) 0.6 0.6 0.6 0.6 0.6 0.6

Substitution uncertainty 0.5 0.5 0.5 2.0 1 .0 2.0

rss 2.9 1 .2 2.9 4.9 1.8 4.9

91
MJTC Primary

Standards

5 - lOVp 5 - 10 mA

20 Hz - 10 kHz

frequency

extension

Special

TE's

5 mA

TE Reference

Standards

5 mA

20 Hz - 100 kHz

TE Reference

Standards

20 A

20 Hz - 50 kHz

TE Working

Standards

5 mA - 20 A

20 Hz - 50 kHz

•build up'

frequency

extension

TVC Reference

Standards

10 V

20 Hz - 100 kHz

TVC Reference

Standards

1-1000 V

20 Hz - 100 kHz

TVC Working

Standards

1-1000 V

20 Hz - 100 kHz

Figure 1. "Road Map"

92
30 Hz 1 kHz

10 kHz

Fi9ure 2. Comparisons of hJTC's

93
50 mA

25 mA

W50E1 W50E2 OF 19

Averacre (50 mA)

-0.7 -0.7 -0.1 -0.7 0.7 0.0

-0.7 -0 8 +0 1 -0.8 0.8 0.0

-0.7 -0 6 0 0 -0.5 +0.5 0.0

+0.7 +0 -1 .0 -0.1 0.1 0.0

-0.1 \ / -0.2

>

0.0 \ / +0.1

+0.4 -0.2 +0.1

-0.7\ +0.1

W25E1 W25E2 NPL14

Average

s- (50 mA)

-1 .0 -0.8 +0.5 -0.6

-0.9 -0.7 +0.U -0.5

-0.7 -0.5 +0.U -0.3

-0.6 -0.3 +0.7 0.0

KEY (for each column) — 50 kHz — 20 kHz — 1 kHz — 20 Hz

10 mA

10 kHz

1 kHz

B10EP2 B10EP5

Average

-0.8 -1 .0 -0.5 -1 .2

-1 .0 -1 .3 -1 .6

-0.8 -1 .0 -o.n -1 .2

-1 .1 -1 .2 -0.2 1 .3

FY 5 mA FZ 5 mA

Average

+0.8 + 1 .0 +0.3 + 1 .2

0 .0 +0.1 +0.1 0.0 +0.2

+0 .2 -0.1 -0.3 -o.u -0.5

+3.2 3.2 3.2

0.0 10 kHz

+0.2 1 kHz

MJTC^

[Primary)

Group,

Figure 3: 50 mA to 5 mA Comparisons

94
95
96
To DC Rev.

Comp.

Set Read

Figure 6. Diagram of EMF Comparator

97
APPENDIX 1

n-COMPENSATION and n-MEASUREMENTS

A. Principle of Conpensation

The basic circuit of the MJTC comparator with the n-compensation

resistor, R^, included is shown in figure 5. Ep represents an adjustable

Lindeck potentiometer.

At the initial balances, with N-0, (N is the reading of the high

impedance nanovoltmeter , nVm), mE^ - E2 - Ep - (The Ep setting is made

by switching nVm in place of R^.) Now if the input voltage Vj^ , is increased

by a small amount, AVj^, then AE^/E^ - n^AV^/Vj^ and AE2/E2 - n2AVj^/Vj^ (from

the definition of n given in section IIIA with Vj^ replacing I). A small

current then flows in the E2-Ep loop. Because of the voltage divider formed

by R2 and Rp, AV2 is less than AE2. Thus if n2 is greater than n^ , R^ can be

adjusted until mAE^ - AV2, maintaining N-0. This makes the steady-state

balance insensitive to small changes in the supply voltages, and makes it

unnecessary to adjust these voltages to keep one of the emf's constant during

an ac-dc determination, as heretofore required.

From the figure and with the output resistance of the potentiometer

included in Rp, AV2 - qAE2t where q - Rp/(Rp + R2). Since the same AV/V is

applied to both TVC's

AE^ AE^ mAE^ n^

TTT" =• TTv- " ^„ . 30 that q

- n^E^ n^E^ qn^mE^ ^ n^

98
B. Baalo TE Equation

Since neither TE has a constant output during a comparison of two TVC's

(or TE's) the development of the comparator equation given in appendix 2 of

C3] is not directly applicable. It will be convenient to use a general

equation for a TVC with slightly different ac and dc Input voltages, and

V^, applied. We may write, as an identity,

,rr. d d ad-*

V = V.[1 + r: + t: J a d^ V. V. ' d d

where V^' is the average of the two directions of dc voltage required to

produce the same emf as Vg^, and is the average of the two directions which

produce an emf, E^.

Then, from the definitions of n and d, and, since - E^j'

,

E - E . V - Vjfl + ^

+ d) , to Ist-order terms.

a d*- nE. '

* d

A similar equation applies for current measurements, with I replacing V.

C. AC-DC CoBparlsona

From the above basic TVC equation, and with the same voltage applied to

both TVC's, as in fig. 5.

1 Id 2 2d

Thus

^1 - ^2 - ^2a ^2d ^la " ^1d

"2^2

99
But - - niE^g and - " "'^Id' ^^^^ polarity shown, so that

^a"^d"^2a"^2d"'"^^1a"^1d^"*'^^2a"^2d^~°'^^1a~^1d^ • Combining these equations,

and noting that q=»n^/n2 and raE^-E2» gives, closely enough,

^-=2 - ^—

i^Te; Sjip-i^

If n^>n2, the potentiometer and Rp are connected across E-j , and the

polarity of the nanovoltmeter is reversed. is adjusted as before, so that

there is no change in N when is increased. Because of the input

resistance, R^, of the divider, analysis shows that

n R R^R^ — " ^s— where R - =—=—

n, R + R, P R +Rj

1 p 1

*^ r d

The comparator formula then becomes

- N

Equations A1B-1 and A1B-2 may be written as

N - N

where the subscript T refers to the TE to which the potentiometer is

connected, and refers to the TE with the lowest emf.

100
D. Precautions In n-Meaaurement

The characteristic, n, is conveniently measured, after the Initial

balance is obtained with the set key down, by changing the Input quantity, Q,

by about 0.1$, noting the change in output emf E, then calculating n=»QAE/EAQ.

The usual precautions in measuring small changes should be observed, as in an ac-dc difference test, to avoid errors from drifts etc.

A more insiduous error can occur, however, if the heater alloy of a TE

has even a small temperature coefficient of resistance, a, so that

R=RQ(1+ae), where 6 is the temperature rise above ambient. Then, since V=IR

for the heater, and 6=kl^

dl dR dl dRdl^ 2R^akIdI

v"l*R'ldIR I R

- ~ (1 + 2ae) since R • R

I o

From the definition of n and the above equation

T^i - Hyd + 2ae).

For the NiCr alloys normally used in low-range TEs, a can be -1x10"^ to

-2x10"^. e is usually about 1.5x10^ at rated I. Thus n^ and n^^ can differ

by 3 to 6$. For alloys like Evanohm the difference is much smaller.

Thus the appropriate n should be measured and used. Since it enters

only in a difference term with this comparator, small errors in determining

it are not significant unless d^-dg is large.

101
E. Square Law Output and RMS Response

It is worth emphasizing that even if n*2, so that E*KI^ with K a constant, a TE inherently responds to the true rms current, I. This can be

shown easily for the simplest case in which all of the heater is at the same constant temperature, 6. Then, equating heat input and output at any

instant, i^R K^e - KgE. R, and K2 are generally somewhat dependent on

e, but if the frequency of the heater current, i, is high enough, thermal

inertia keeps 6, and hence E constant. Then term by term integration with

respect to time, t, over one period, T, and division by T, gives, directly,

I R-K2E. Because of the temperature dependence of R and K2 the output is not

exactly proportional to 1^, but it is still a measure of I, the rms current.

This can also be shown to be true under far more general conditions (see

appendix 3 of [1]).

102
Appendix 2

EMI, Harmonica and DC Offset

A. Test for Line-Borne Electromagnetic Interference (EMI)

Only about 1 of heater power, over a wide range of frequency, can produce 0.1$ of rated emf in a typical 5-tnA TE. Thus TE's are sensitive to

EMI, particularly at TV and FM frequencies. (They were the detectors in

early field-strength meters.) Laboratory leads at these frequencies can transfer radiated energy fairly efficiently to the heaters. Of course if the

leads are twisted or shielded they make very poor antennas. However

shielding is not completely effective, and in areas of high field strength

(up to 1 V/m in some areas) errors can result if the TE responds to currents

at both the main and interfering frequency.

However, because of their square-law characteristics (E-KI^ where I is

the rms current) a small heater current, Ig, at a different frequency than

the main current, Ijjj, does not cause as large an error as the ratio a-Ig/Ij^

might indicate.

If the response sensitivity is the same at both frequencies

E - Kl2 = KIjjj2(l + a^]

Without Ig, Ej^ - KIjQ^. Thus [e-E^}/E^ - ^^/Ej^ - a^. With only Ig present,

Eg - Klg^ so that Eg/Eni - a^.

If a measurement of Ij„ is desired, AE/E^ is an error term. It can be

evaluated to see if it is significant by making a measurement with all of the

connections in place but with the power supplies producing Ijjj turned off.

103
In the more general case K is somewhat dependent in E, so that the

sensitivity of this test is affected slightly. Also, if both TE's in an ac-dc comparison respond equally well to currents at both frequencies, there

will be no EMI error. However the EMI test will determine this also.

In the MJTC comparator, RF bypass capacitors (of very low inductance)

are connected from each ungrounded lead to the metal tank to reduce the

possibility of EMI errors. The tank itself is an excellent shield.

B. Effect of Haraonlcs (see Section VII)

Small amounts of low-order harmonics in the input voltage generally

cause insignificant errors in the measurement of the rms value by a TVC, or

in the comparison of two TVC's. This is a great advantage for true

rms-raeasuring instruments. This can be shown rather simply for a wave having

a fundamental voltage, , and a single harmonic, V^. Then V»(v^^+Vj^^)^^^.

Because of reactance, skin effect, etc., the response of the TVC may be

different at the two frequencies, so that E-kV^-k^V^+kj^V^^. Let Vj^/V^-q and

K^^ - K^(l + e^). Then E - k^V^ll + Q^®h^ ^^"'^ proportional error is

only q2e^.

For several harmonics this becomes Z Qm^ejnt where m refers to the order

of the harmonic.

C. DC Offset

The above equations should apply even if the harmonic is of zero order

(dc). However if the TE has a dc reversal difference (which can be caused by

Thomson or Peltier heating) there is an added output, E-KQVQg, proportional

to the first power of any dc offset voltage, V^g, in the ac wave, V^. Then

E - k(v 2 V 2)1/2 ^ K V - KV^d *

^ a OS ^ o OS E

104
With a dc voltage, v^j, equal to V applied and then reversed, the dc

reversal difference in output is AE - 2KqV. Thus

E = kV^d . 8H)

where

V

AE . 03

g - — and u =

Note that this is proportional to the 1st power of u, and that, for a square

law TE, g is twice the value of the proportional dc reversal difference as normally defined (AV/V for constant E).

105
Appendix 3

Effect of Relay Off-Time

Let the response, E, of a TE be approximated by a simple exponential

with a time constant, t^. Then after the steady-state heater current is

turned off for a time t^KKt^ (the off-time of the relay when switched), we have

E - E^e . Ej,(l - t^/tj .

Thus AE = Ej.-E = Ef^t^/t^.

When the relay is closed (with the same heater current), E recovers to

Ej. exponentially, so that

-t /t

E = E^ - AE e ^

°

where tg = t-tQ. Then p =[Ef-E]/Ef = b/e^ where b » AE/Ej. = ^o^^c

a = tg/t^ » t/t^ when t3>>tQ. If p = 1x10*^ (recovery to 1 ppm) , t^ =» 4x10 s, and tQ=2 s, then b = 2x10 and a»5.3. Thus t - at^, = 10.4 s.

Interestingly, when t^ = 4x10~^ s, t « 15.2 s to recover to 1 ppm, only

a modest increase for this 10-fold increase in off time.

106
Appendix 4

Effects of Bead Heating

A. Relative Effectiveness of Bead and Heater Power

Widdls has shown (equation 17 of [19]) that the midpoint temperature

rise, e„j, of the heater of a TE carrying a current, I, taking into account

the heat loss through the thermocouple wires as well as the heater, to heat

sinks at the ends (all at 9=0) is 6^ » DJ,^/2M. Here the very small Peltier

and Thomson heating is neglected, D - I'-p/k K., p and K are the electrical

resistivity and thermal conductivity of the heater, A is the cross sectional

area and 21 the length of the heater, and M - 1+K^/Kq, where is the sum of

the thermal conductances of the two thermocouple wires in parallel and Kq is

the sum of the conductances of the two halves of the heater.

Since kg - 2KA/II and 2pJ,/A - R^^, the heater resistance, we find that

^m ' 2(Kq+K^)

where

"h - ^\

If the bead between the heater and the hot Junction of the thermocuple

has a resistance R^, its power dissipation is P^-I^ "b» The added

temperature rise from this small heating is AS - Pjj/(K-| +Kq) . Thus bead power

is twice as effective as heater power in raising the temperature of the hot

junction.

107
B. AC-DC Difference Caused by Bead Heating

One end of the heater of a TVC is grounded, and the thermocouple circuit

is usually grounded also for safety and to keep the bead voltage low. Thus

half the heater voltage is across the bead. From the formula in section V-B

the significant current is through the ungrounded heater terminal. The

heater power is, to first order terms,

2 2

. . il^ (1 - • - a) h 2

where

a , - « 1 ^1

The temperature rise, with the effect of bead power included, is

e - UP^+ 2UP. - UI,^R. (1-a) + 2Ua^I,^R^ h b 1 h lb

where

Thus

U - 1/2(k + kj

o 1

e - UI.^R. (1 - a + 2a\ /R. ) - UI.^R.

1 h b h 1 h

with no error from the bead current, to first order terms.

This will, of course, not be true if neither heater terminal is at

ground.

108
C. Two TE'3 in Series

A similar but more involved analysis shows that if two equal

series-connected TE's are used in a grounded TVC, the resultant output will

also be unaffected by the bead currents.

)

109
Appendix 5

Impedances and Substitution Corrections

Ingles has studied the general Impedance relationships in TV's and the

effects of changing TE's.[9] However the following alternative approach,

with its explicit treatment of phase defect angles, was developed at NBS. It

has been used very effectively for several purposes, and so is given here.

It is analogous to a much earlier unpublished NBS treatment of shunted TE's

for current measurements.

A. Inpedance of a TVC

A TVC set with well-defined ac-dc characteristics (independent of the

surroundings) generally consists of a shielded TE in series with one of a group of three-terminal shielded resistors, as shown in fig. 1 of [2]. As

shown in section VI of [2], the ac-dc difference of a combination as a TVC

is, in the present notation, d^ - d^+d^,, where d^ - {z^^-R^]/R^ and

^c " Uha-Ihd)/I In these expressions, - Vj^/Ij^, and is the magnitude

of the impedance of the combination, with an applied rms voltage and a resultant heater current Ij^. 1^^^ and Ij^^ are the magnitudes of the ac and dc

currents required to produce the same output emf.

This is a very general equation, applicable for any series resistor and

TE combination. To evaluate the component impedances we may write as an identity

Z.t^i- (V, - V. )/I. + V. /I. » Z + Z. h''h hh s h

where V^ is the voltage across the heater. The bars signify either phasors

or complex numbers, and their absence signifies magnitudes.

110
We may also write

Z /R - 1 + a + jb and Z. /R. - 1 + c + 1e

3 3 n n

with a, b, c and e each much less than unity for a useful TVC. Rg and R^i the dc values. Then

h .

1 ^ a ^ Jb ^ p(1 ^ c ^ Je) . ^ g ^

R^. 1 + P

' J"

where p - R^/Rst S (a + pc)/(l + p) and h - (b + pe)/(1 + p). Note that

h - tan • the phase defect angle in radians, closely enough, since g

and h are very small. Then to a sufficient approximation

|i . [1 * 2g * h2]^^2 . , , g V

and

vnien p ^ 1, h^/2 < 1 x 10"^ if b and e are each less than 1 mrad, so that the phase angle term is negligible. In this case

^t "8. ^3 - a, and d^ - c

where dg - (Zg/Rg) - 1, d^ - (Z^/R^) - 1.

Ill
B. Substitution Corrections (See Section IX)

Consider two different TE's of the same nominal Rj^, each with a known d^

as a current converter, and a known d^^ as a voltage converter (with no series resistor). The difference in d^ of a TVC range with the same series

resistor, is, if all phase angles are negligible,

dv2 " dvl - g2 - 81 + - del - p(c2 - Q'\)/i^ + p) + <ic2 " .

But

p/(1+p) - Rj^/R^ and C2 - c^ - d^a - d^i • (dvh2 " ^c2^ ~ (dyhl " )

.

Then

^2 " ^1 " '^h2 ^h1 J * t^c2 ^c1 ^

•

Thus the substitution correction for any range is expressed in terms of

the directly measurable d's of the new and old TE's as current and voltage

converters

.

If the phase angles are not negligible a term, h-|2-h2^, must be added.

This reduces to

pb(e^-e^) ^ 2p^(e/ - e^^)

h - h - ^

. (1 + p)"^

Thus it is zero if e2 » e^

.

If b = 1x1 » e^ and 62 - 2x1 (as rather extreme examples) this

added phase angle correction is less than 1 ppm even at p»1 (low voltage

112
range). It Is much smaller for higher ranges, even If b Is larger (as it may

be because of capacitive currents).

At NBS a commercial impedance comparator has been very convenient for

comparing phase angles to better than 1 mrad up to 100 kHz. It is used to

screen the series resistors and TE heaters to make sure that their phase

angles are negligible.

C. Interchsuiged Series Resistors

As explained in section V, two nearly identical 30 V 3-terminal

resistors, and of negligible phase angles, in separate coaxial

shields, are connected in series with two nearly identical TE's. The

resulting TVC's are compared, to evaluate dQ^-dQ2 of the TE's as current

converters. Then the resistors are interchanged and the comparison is

repeated to essentially eliminate inequalities in the magnitudes of the

reactances of the resistors. For the first measurement, with in series

with TE-| and R2 in series with TE2f

a^ + pc^ - (ag + pCg)

+ d - d

1 + P cl

For the second measurement, with and R2 interchanged

a- + PC, - (a, + pc-)

c^) + d

c1 - d - d

cl

d closely enough,

since

-E h

. i

1+p ' R^

^

for the usual 0.5 V TE's.

113
A more extended treatment, including the effects of phase angles, leads

to added terms in , and e2, similar to those given in appendix 5B.

They are negligible if they do not exceed the limits given there.

D. AC-DC Difference of Series Resistor

For a given range of a TVC, from the equations of appendix 5A, with all

phase angles negligible,

d + pd.

d = d. + d » ; + d

V t c 1+p c

Thus

R. Rj,

d = (1+p)(d -d ) - pd^. = ~ (d -d ) - d. s *^ vc *^h R vc R h

s s

since d^ = - d^, where d^j^ is the ac-dc difference of the TE as a TVC.

E. Effect of a Small Phase Difference Between the Paralleled Impedances in

the Current Build Up

The current through the single higher-range TE in the build-up is the

phasor sum of the currents through the two paralleled lower-range TEs. It

will differ from the sum of the magnitudes of the two currents, I^ and if

the two are not in phase.

For simplicity let - I^/O, I2 - 12-^ " with e«1 (in radians).

The bar signifies a phasor quantity, and its absence a scalar. Since is

the reference,

1 = 1^ ^2 " '1'^ * "

"""I (cose+jsine)

= I. [(2 + 6 /2) + je] (neglecting higher order terms in e)

114
From this and by continuing to neglect terms in and higher

I - 21^(1 - 8^/8)

The proportional difference is

e - (21^ - I)/I » .

If e » 3x1 0~^, 6 « 0.1x10"^ radians. Since the two impedances, Z, and are in parallel,

This will be true only if Zg - Z^/.-0. Thus the phase difference between

and Z2 must be less than IxlO"^ rad to keep the phase error less than about

0.3 ppm.

115
Appendix 6

RECOMMENDATION

of the Working Group on the Statement of Uncertainties

presented to Comite International des Poida et Mesures

Assignment of experimental uncertainties

RECOMMENDATION INC-1 (1980)

1 . The uncertainty in the result of a measurement generally consists of

several components which may be grouped into two categories according to the

way in which their numerical value is estimated:

A - those which are evaluated by statistical methods,

B - those which are evaluated by other means.

There is not always a simple correspondence between the classification

into categories A or B and the previously used classification into "random"

and "systematic" uncertainties. The term "systematic uncertainty" can be

misleading and should be avoided.

Any detailed report of the uncertainty should consist of a complete list

of the components, specifying for each the method used to obtain its numerical value.

2. The components in category A are characterized by the estimated

variances, a^^, (or the estimated "standard deviations" 3^) and the number of

degrees of freedom, \,^. Where appropriate, the estimated covariances should

be given.

3. The components in category B should be characterized by quantities Uj^,

which may be considered as approximations to the corresjxjnding variances, the

existence of which is assumed. The quantities u^^ may be treated like

116
variances and the quantities Uj like standard deviations. Where appropriate,

the covariances should be treated in a similar way.

The combined uncertainty should be characterized by the numerical value

obtained by applying the usual method for the combination of variances. The

combined uncertainty and its components should be expressed in the form of

"standard deviations".

5. If, for particular applications, it is necessary to multiply the com- bined uncertainty by a factor to obtain an overall uncertainty, the multiplying factor used must always be stated.

117
300 Section 5 Multijunction Thermal Converters as the NBS

Primary AC-DC Transfer Standards for AC Current

and Voltage Measurements

FRANCIS L. HERMACH, life fellow, ieee, JOSEPH R. KINARD, senior member, ieee,

AND JOHN R. HASTINGS, member, ieee

Abstract—k set of multijunction thermal converters (MJTC's) has been established as the NBS primary ac-dc transfer standard for ac current and voltage measurements. This choice is based upon the ad- vantages of the MJTC's, the development of an improved EMF comparator, an extensive series of intercomparisons, and numerous supporting investigations.

I. Introduction

THE USE OF ac-dc transfer standards enables ac currents and voltages to be accurately measured in terms

of their dc equivalents and the underlying basic electrical

units. Although electrostatic and electrodynamic instru- ments can be used for ac-dc transfer, electrothermal in- struments such as thermal converters are generally used

because of their greater accuracy and wider current, volt- age, and frequency ranges. The ac-dc difference of a transfer standard is conventionally defined as 6 = ( 2a ~

Qd ) I Qd- where 2<y is the average of the two polarities of

the dc input quantity (current or voltage) required to pro- duce the same output as that produced by the ac input Q^.

Hence (2« = (2<y ( 1 + 5).

Thermal voltage converters (TVC's) and thermal current converters (TCC's) are calibrated at NBS by comparing them with transfer standards whose ac-dc differ- ences have been carefully determined. Most thermal

converters contain a single-junction thermoelement

(SJTE), consisting essentially of a wire heated by the ac

or dc current and a thermocouple to sense the temperature

rise. A multijunction thermal converter (MJTC) can have

up to 200 or more thermocouple junctions uniformly

spaced along a bifilar heater to increase the output EMF

and reduce thermoelectric errors [1]. An investigation of

MJTC's at NBS several years ago revealed a number of

shortcomings which were overcome by changes in design

or by compensation. The average ac-dc difference of a

set of eight MJTC's of different construction and current ranges and from different manufacturers was shown to be

very probably 0.5 ppm or less from 30 Hz to 10 kHz [2]. These MJTC's have recently been reevaluated in a study

Manuscript received June 23, 1986. This work was supported in part by

the Calibration Coordination Group of the U.S. Department of Defense. The authors are with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899. IEEE Log Number 8613427.

of the uncertainties of the NBS standard TVC's and TCC's

[3], which was undertaken partly in response to the need

for better ac-dc transfer standards to support improved ac

calibrators and voltmeters. Four additional MJTC's of two new designs have also been compared to the NBS set.

II. Precision Comparator System

The comparator system, consisting of an air bath, an EMF comparator, ac and dc supplies, and a protective

circuit, is shown in simplified form in Fig. 1, with an MJTC and an SJTE connected in parallel as voltage converters. A more detailed circuit diagram of the comparator itself is given in [3]. The thermal converters are lo- cated in the air bath, which is a thermally lagged metal

tank. The comparator basically works as follows. With a

resistive divider a fraction of the higher output EMF of one converter is opposed to the full output EMF of the

other. The three-stage divider mV\ and the Lindeck po- tentiometer Ep are first adjusted to bring the detector, a commercial chopper-type nanovoltmeter (nVM), near zero

for both positions of the key with dc applied and are then

left undisturbed during an ac-dc determination. Changes

in the balance are measured sequentially as ac, each polarity of dc, and then ac are applied at 40-s time intervals. An operational-amplifier integrator and timer are used to integrate the 1-V output of the nVM for 10 s and display

the average on a digital voltmeter. This significantly re- duces the uncertainty from Johnson and nVM noise and

simplifies the data collection. Typically, repeated 10-s averages fluctuate by less than one-tenth of the peak-topeak noise seen in 10 s on the nVM, which is about 35

nV in the comparison of two MJTC's, approximately 3.5

times the calculated Johnson noise. To provide the best

compromise between loading and noise level, the input

resistance of the divider is switched from 10 kfi in the comparison of two MJTC's or an MJTC versus an SJTE

to only 1 kfi (with one stage bypassed) in the comparison

of two SITE'S. A reversal compensator (RC), consisting

of an adjustable 0-13-/xV dc source with negligible change on reversal [2], is used when the dc reversal difference is larger than the nVM range. A study of protective circuits to guard the converter

heaters against burnout showed that a simple arrangement

U.S. Government work not protected by U.S. copyright

118
HERMACH ei ai: MULTIJUNCTION THERMAL CONVERTERS AS NBS AC-DC TRANSFER STANDARDS 301

Fig. 1. Precision comparator system with an MJTC and an SJTE connected as voltage converters.

of a pair of back-to-back Zener diodes directly in parallel

with the heaters and an expendable thermoelement (TE)

in series with the combination provides adequate protection against reasonable kinds of overvoltages. The Zener

voltages are chosen to be about twice the supply voltages.

TE's with large dc reversal differences, which are rela- tively inexpensive, serve well as fuses. The converters require time to recover following any

interruption in the heater current. The resulting moderate

restrictions on the off times of the input relays, so that the converters can recover sufficiently within 30 s after switching, are discussed in Appendix C. Relays with an OFF time of less than 0.5 ms are used to switch between

the ac and dc inputs.

When the two converters being compared have equal

values of n, where n = li^E / EAI, I is the heater current,

and E is the output EMF, the steady-state balance is in- dependent of small changes in the output of the ac and dc

supplies. This reduces the uncertainty in the measurement

and the need to keep the EMF of one TE constant during

a determination. Although n is very nearly 2.0 for MJTC's, it ranges from about 1.6 to 2.0 for the SJTE's

in the reference standards against which the MJTC's must

be compared, and thus some of these advantages are lost. To compensate for the difference in n values, an adjustable resistor R„ was added to the comparator circuit, as shown in Fig. 1. If the input voltage changes slightly, a small current will flow in the E\-Ep loop. Because R\ has

a finite resistance and forms a divider with R„, the change

in voltage across the three-stage divider will be less than

the change in E\ . The compensating adjustment of R„ is discussed in Appendix A as part of the derivation of the

basic equation for comparing two TE's:

where subscript t refers to the TE with the higher n, to which the Lindeck potentiometer is connected, and subscript 5 to the other. A^^ and are the average readings

of the nVM with ac and dc applied, respectively, and

is the lower of the two TE EMF's. If the TE on the right

side of Fig. 1 has the higher n, both Ep and R„ arc switched

across E2 and the polarity of the nVM is reversed. The

basic equation then remains the same. If is in nanovohs

and E in millivolts, A is in parts per million. In a recent paper [4] Inglis has shown that the conventional definition of ac-dc difference is in error if the converter has a dc reversal difference q = \ Q\ ~ Qi] / Qj,

with £, = E2. Here subscripts 1 and 2 refer to the two

polarities. He has proposed a new definition, 8' = ( Q,, -

Q'd ) / Q'd^ where Q'^ is the dc input that, when applied with

the same magnitude in both positive and negative polarities, produces a mean output (E, + Ei)l2 equal to the ac output Ea- It is free of this error but can be awkward

to use directly. The difference between the two definitions

depends on a number of factors. However, for a square- law TE in which q is approximately independent of £, the difference is only about 2>q^/%. If ^ = 5 x 10""* (rather

large for a TE intended for precision measurements), the difference is less than 0.1 ppm. For the NBS primary

MJTC's the values of q are less than 1 x lO""*, and that makes the difference between definitions insignificantly

small. The results given in this paper were evaluated with

the conventional definition of b. To test for possible errors caused by electromagnetic

interference (EMI), "phantom" measurements were made

with the supplies connected to the TE's but switched off, as discussed in Appendix B. No such errors were found.

As a precaution all ground leads and shields are connected

to the grounded metal tank, which is an excellent shield,

and all ungrounded leads are bypassed to the tank with

capacitors (of very low inductance). The effects of har- monics and dc offset in the ac input are also discussed in Appendix B.

III. Results

The improved EMF comparator has been used to inter- compare very thoroughly four of the eight MJTC's that were investigated earlier [2]. These four are labeled A50-

14, BlO-44, B5-2, and ClO-1 and will be called the basic

subset. Five other MJTC's (labeled D-55, D-60, E-72,1,

E-73,1, and BlO-2) have also been studied. The results of the intercomparisons as voltage converters at 30 Hz, 1 kHz, and 10 kHz are shown in Fig. 2(a)-(c), respectively.

The number near the head of each arrow is the measured

b, — Sj in parts per million, where b, and 6j are the ac-dc

differences of the MJTC's at the head and tail of each

arrow, respectively. The measurements within the basic

subset had a pooled standard deviation of the mean of 0. 16

ppm. There was no evidence that the diflFerences between

averages in repeat tests were statistically significant.

To determine the values assigned to each MJTC in Fig.

2, the best estimate of 6, - b^ for CI 0-1 and B 10-44 was

first obtained from the three paths between them; double

weight was given to the direct path. The best estimates of

6 for the four basic MJTC's, on the basis that their aver- age is zero at each frequency, were then easily determined

and are shown to the nearest 0. 1 ppm in each ellipse. A50-

119
302 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-36, NO 2, JUNE 1987

(a)

(b)

-0.4^/Rin.9\

-0.8/ -0.7 V+0.2y V-0.2/

-0.4 /cio-n -0.05 /B10-44\ + 0.6 \-o^

V+0.1

,

+ 0.4

(c)

Fig. 2. Results of the intercomparisons of M,ITC"s as voltage converters

at (a) 30 Hz, (b) 1 kHz. and (c) 10 kHz. At 30 Hz the MJTC's outside

the dashed line are compensated; those inside the dashed line are uncompensated.

14 is not useful at 10 kHz because it has magnetic leads. BlO-2 was not included in the basic subset because it was

compared only to B5-2 in early measurements which were

not repeated. D-55, D-60, E-72,1, and E-73,1 are newer MJTC's which have been compared more recently.

The values of 8 for the four basic MJTC's and for B 10- 2 differ from the corresponding values obtained in the pre- vious investigation by no more than 0.4 ppm. At 1 kHz,

where the previous values for all eight were no larger than + 0.2 ppm, four of the five remeasured MJTC's have the same value as before and the other (B5-2) is 0.1 ppm

lower. This excellent agreement demonstrates that these

MJTC's of different design and construction are very sta- ble.Comparisons as current converters have also been made

between primary MJTC's and specially constructed ref- erence-standard SJTE's. The results of the intercompari- sons between a MJTC of one design, F50-19, at 50 mA

and another MJTC of a different design, ClO-1, at 5 mA,

through intervening steps of SJTE's, are shown in Fig. 3. Two almost identical TE's of one range were first compared in series at rated current (for example, FZ and FY

at 5 mA). They were then connected in parallel to form a

unit that was compared in series with each of two TE's of

the next higher range (B10EP2 and B10EP5 at 10 mA),

and so on. This is the method used in [5]. As in that work,

pairs of almost identical low-reactance resistors were connected in series with the heaters to ensure that the paralleled currents were sufficiently in phase. The phase angles

of the two paths must be equal to better than 1 mrad, and

the two paralleled TE's must be well matched [3], [5]. This method has the advantage that each TE is used at or near rated current, so the results are not affected by

current-dependent errors. The 50-mA MJTC was as- signed midfrequency 6 values of zero, based on theoretical [1] and experimental [2] analyses. The numbers to the

right inside the squares are the 6 values determined from

the intercomparisons at 1 and 20 kHz with forced closure

of each measurement triangle. The closure adjustments of

the sides of the triangles averaged only 0.1 ppm. Results

of the individual comparisons and at other frequencies (20

Hz and 50 kHz) have been omitted for simplicity. Assignment began at the 50-mA level and progressed down

to the 5-mA FZ and FY TE's. At 5 mA, FY was compared

to another MJTC, ClO-1, with the results shown on the

left of the square. The values from the build-down pro- cess and those by direct measurement at 5 mA agree to within 0.3 ppm. Thus the systematic errors in the chain

are very small. The pooled standard deviation of the mean

of four determinations in an evaluation of 8, — 8, was 0.28

ppm.

Of particular interest is the excellent agreement at 1 kHz

(within 0.6 ppm) between the MJTC's and the averages

of the two 50-mA and the two 25-mA TE's. Each of these

four single-junction converters has an Evanohm heater and

Evanohm stems (lead-in wires) between the heater and the

glass press (which serves as a good heat sink). Thus the

Peltier as well as the Thomson errors of each TE should

be very small. Any residuals would probably not be the same in all of them, nor the same as those in the MJTC's.

This agreement provides valuable additional confirmation

that the ac-dc differences due to thermoelectric effects in the NBS standards are considerably less than 1 ppm.

IV. Significance of Midfrequency Evaluation

Low-frequency ac-dc diflFerence effects, due to doublefrequency ripple, and high-frequency effects, due to re- actance and dielectric loss, are generally negligible over

a rather broad midfrequency range. This leaves only what

have been called "intrinsic errors," caused chiefly by

thermoelectric efl'ects in the heaters themselves. Wilkins

showed theoretically [1] that these can be less than 0.1

ppm with attainable constructions of MJTC's, which is much less than for SJTE's. Nevertheless, imperfections

in materials and heat treatment can cause actual MJTC's

to differ substantially from theoretical models, so that the experimental verification outlined above is very important

[2].The agreement shown in Figs. 2 and 3 is thus highly

significant. It is very improbable that any errors would be

the same for such a diverse set of thermal converters. Supplemental studies showed no significant effects from

known sources of ac-dc difference at 1 kHz.

Zhang and Zhang have shown [6] that MJTC's have an ac-dc difference caused by the Thomson effect in the re- gions near the ends of the heater. For an experimental

MJTC with a constantan heater and 44 thermocouples they

120
HERMACH ei ai. MULTUUNCTION THERMAL CONVERTERS AS NBS AC DC TRANSFER STANDARDS 303

SidA 10mA 25mA 50mA

10kHz 1kHz

SJTEs

MJTCs

Fig. 3. Results of the intercomparisons of MJTCs and SJTE's as current converters between 5 and 50 mA. The numbers to the right inside the squares are the 6 values determined at 1 and 20 kHz.

found that the ac-dc difference 6^ resulting from the end

effects was about 10 ppm at three times the normal heater

current. Since 5^ is a second-order effect, it would be ex- pected [7] to be proportional to the square of the Thomson

coefficient a, and hence less than 0.1 ppm for similar

MJTCs with Evanohm or comparable heaters for which

a is only about 5 percent as large as it is for constantan.

In addition, bg would be expected to become smaller as

the number of couples is increased and as the temperature

rise is reduced. The NBS MJTCs have Evanohm or comparable heaters with approximately 50-200 couples and

operate at temperature rises of only 15-40°C at rated current. The excellent agreement among them is good evi- dence that end effects are negligible.

V. Low-Frequency Evaluation

As the frequency is reduced and the heater temperature

begins to follow the double-frequency variation in instan- taneous power, ac-dc difference effects are caused by the

temperature dependence of the electrical resistance and

thermal conductance. Measurements have shown [2] that these effects are much more complex in MJTCs than in SJTE's, and no satisfactory theory is known to us. Effects of 3 ppm or more have been observed at 30 Hz in other- wise satisfactory 5-mA MJTCs. The major source is apparently a change in the effective heater resistance [2]. Fortunately, MJTCs can be compensated by adding a matched resistor (in series for voltage, in parallel for current) whose resistance is approximately equal to the heater

resistance [2]. In Fig. 2 at 30 Hz the MJTCs with the

letters A, B, or C '\r\ their label are compensated in this way, and their values of 6 agree with the average of the subset to within 0.3 ppm. The newer D and E MJTCs

are uncompensated, but even so, their 6 values are between 0 and — 1.2 ppm at this frequency.

VI. High-Frequency Evaluation

Theoretically, MJTCs can have ac-dc differences that are smaller than 2 ppm up to 100 kHz [1], but TVC in- tercomparisons have shown much larger differences in the NBS primary set, whose use is presently limited to no more than 10 kHz. This limit has been extended to 100

kHz (with poorer accuracy) by complex intercomparisons

of reference TVCs, each consisting of an SJTE in series with a resistor [3]. Additional measurements of newer

types of TVCs and MJTCs are in progress, as are further

studies of the causes of ac-dc difference.

VII. Summary

The accuracy and the stability of the NBS set of MJTCs

have been verified. The average ac-dc difference of this set is still 0.5 ppm or less from 30 Hz to 10 kHz. An

improved comparator system has been used to make precise intercomparisons of these MJTCs and the SJTE's in the NBS reference standards. As a result of this work these

MJTCs now serve as the NBS primary ac-dc transfer standards for ac current and voltage measurements.

Appendix A

Comparator Equations and aj-Measurements

A. Comparator Equations

Since neither TE has a constant output during a comparison of two TVCs (or TE's), the development of the comparator equations given in [8, Appendix II] is not di- rectly applicable to the improved comparator. It will be

convenient to use a general equation for a TVC with

slightly different ac and dc input voltages, and V^, applied. We may write, as an identity

121
304 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. lM-36, NO. 2, JUNE 1987

where is the average of the two polarities of dc voltage

required to produce the same EMF as V^, and is the average of the two polarities that produce an EMF, fj.

Then from the definitions of n and 6 given in the text and

the fact that = Ey.

En E^

1 + ^ + 5 nE^to first-order terms.

(A2)

A similar equation with / replacing V applies for current measurements.

The basic circuit of the comparator, with the n-compensation resistor /?„ included, is shown in Fig. 1. At the

initial balances, with N = 0 (N is the reading of the highimpedance nVM), = Ej and ^'| = Ep, where is the voltage at the input of the divider and Ep refers to the Lindeck potentiometer. Now if the TVC input voltage Vi changes by a small amount A K, , then

and

AE2 AK,

17 = "'^ (A3)

from the definition of n. Note that second-order terms are neglected in this approximation.

The small change in the input voltage causes a small

change A/, in the output current from the MJTC, so that A K| = A^i — /?, A/|. By the superposition theorem

AV, = AE, 1 pAE\ :a4)

where p = Rp/{R, + Rp), Rp = R^Rd/iRn + Rd), and

Rj is the input resistance of the divider. If R„ (which in- cludes the output resistance of the potentiometer) is adjusted so that AA^ = 0

AE2 = mAVi = mpAEi.

Then from (A3) and (A5)

AE2 A£, AE2 SAE2

n2£^2 ^i^^i mpn\E\ pn\E2

(A5)

(A6)

since E2 = msE\ at balance, where s = Rj/iR^ + Rj).

Thus

P =

5«2 (A7)

Note that 5«2 must be less than in this configuration.

In an ac-dc test the same voltages are applied to both

TVC's, so that from (A2)

6, =

Ela - E2d \d

njE^d

Since (A4) is a linear equation

(A8)

Na-N, = E2a - E2d ' ^ ( F„ - K„ ) = Eia - E2d - mp{E\^ - E\j) (A9)

where the subscripts a and d correspond to the ac and dc

input voltages. Thus

. s - E2d rns{E\^ - E\j)

0] — 02 = —

n2E2d n\E2d _ {Eia - Eld) ~ fnpiE^^ - E\j)

n2E2d

niEId (AlO)

If ^2 > both Ep and ^„ are switched across E2 instead

of £| and the polarity of the nVM is reversed. /?„ is then

adjusted as discussed above, so the remains the same when K, changes. In this case a similar (and simpler) analysis shows that

6, =

A^^ - Nn

(All)

n\E2d

More generally, (AlO) and (All) may be written as

b, - b. nsEi

(A12)

where b, refers to the TE or MJTC to which the Lindeck

potentiometer is connected and £/ is the lower of the two

EMF's.

B. Precautions in n-Measurement

The characteristic n is conveniently measured, after balance has been obtained, by changing the input quantity

Q by about 0. 1 percent, noting the change in E, and then

calculating n = QAE/ EAQ. The usual precautions in measuring small changes should be observed, as in an acdc difference test, to avoid errors from drifts, etc. A more insidious error can occur, however, if the heater

alloy of a TE has even a small temperature coefficient of

resistance a, so that R = Rq{\ + ad), where 6 is the

temperature rise above ambient. Then since V = IR for

the heater and d = kI^ dV__dI dR_d[ dRdl dl IRpaKldl

'v~I^R~I^dIR~l'^ R

= y ( 1 -I- lad) since Rq ~ R.

From the definition of n and the above equation

«, = /7„(1 + lad).

For the NiCr alloys generally used in low-range TE's, a is typically around -1 x 10""* to -2 x lO""*. 6 is usually about 1.5 x 10^ at rated current. Thus n,, and n, can differ by 3-6 percent. For alloys like Evanohm the

difference is much smaller.

The appropriate n should be measured and used. Since

122
HERMACH CI al.: MULTIJUNCTION THERMAL CONVERTERS AS NBS AC-DC TRANSFER STANDARDS 305

it enters only in a difference term for this comparator,

small errors in determining it are not significant unless 5, — 6, is large.

Appendix B

EMI, Harmonics, and DC Offset

A. Test for Line-Borne Electromagnetic Interference

(EMI)

As little as 1 /nW of heater power, over a wide fre- quency range, can produce about 0.1 percent of rated EMF in a typical 5-mA TE. Hence TE's are sensitive to EMI, particularly at TV and FM frequencies (they were

the detectors in early field-strength meters). Laboratory

leads at these frequencies can transfer radiated energy

fairly efficiently to the heaters unless, of course, the leads

are twisted or shielded. Shielding is not completely effective though, and in areas of high field strength (up to 1 V/m) errors can result if the TE responds to currents at both the interfering and the measurement frequencies.

However, because of their square-law characteristics {E — KI^, where / is the rms current), a small heater current

at a frequency different from that of the measurement

current does not cause as large an error as the ratio w

= Ig/Im might suggest. Here

i = {ii + iif' = iM +

^'y^'.

If the response sensitivity is the same at both frequencies

E = KI^ = Kli{\ + w^).

Without 4, the output is simply E„ = Kll,. Thus (E -

E„) / E^ = AE/ E„ = w^. With only 4 present, the out- put is Eg = Kll and EjE^ = w^.

If a measurement of I„ is desired, AE/ E„ is an error term. It can be evaluated to see if it is significantly by

making a test measurement with all of the connections in place but with the power supplies that produce I„ turned

off. In general, K is somewhat current dependent, which

slightly affects the sensitivity of this test. There will be no EMI error if the TE's in an ac-dc comparison respond

equally well to the currents at both frequencies. The EMI

test will determine this too.

B. Effect of Harmonics

Small amounts of low-order harmonics in the input

voltage generally cause insignificant errors in the measurement of the rms value by a TVC or in the comparison

of two TVC's. This is a great advantage for true rms- measuring instruments. It can be shown rather simply for

a wave having a fundamental voltage and a single har- monic Vf,. Then V = (V] + Vl)^^^\ Because of reactance, skin effect, etc., the response of the TVC may be

different at the two frequencies, so that E = kV^ =

k\ y] + vi. Let VJV^ = r and k^ = k,{ \ + e^). Then

E — k\ 1 -f- r^e,,), and the proportional error is only

r^Cf,. For several harmonics this becomes E rl,e„, where

m refers to the order of the harmonic.

C. DC Offset The above equations should apply even if the harmonic

is of zero order (dc). However, if the TE has a dc reversal difference caused by Thomson or Peltier heating, there is an added output E„ = K„V,„, proportional to the first power of any dc offset voltage V,,^ in the ac wave V^. Then

if E„ « E

E = k{Vl + Vi) + K„V„s = kV'(^\ +

With a dc voltage equal to V initially applied and then

reversed, the dc reversal difference in the output is Af =

2K„ V. Thus

where

AE

and

Note that this is proportional to the first power of u, and

that, for a square-law TE, g is twice the value of the dc

reversal difference as usually defined (A K/ Kfor constant E).

Appendix C

Effect of Relay Off Time

Let the response of a TE, whose final value is Ej, be

approximated by a simple exponential with a time constant t^. When the steady-state heater current is inter- rupted for a short time tg « t^, where t„ is the off time

of the relay when it is switched, we have

E = Efe-'"/'^ ^ Ef{\ -tjt,).

Thus AE = Ef - E = Eftjt,.

After the relay is closed again (and the same heater current is restored), E recovers to £y exponentially, so that E = Ef- A £e

where t, = t - t„. Then h = {Ef - E) j Ef = h j e" , h = AE/Ef = t„/tc, and a = t,/t^, which approaches t/t,. when t; » t„. If we assume that = 1 x 10~^ (recovery

to 0.1 ppm), = 4 X 10"" s, and t, = 2 s, then b = 2 X 10"'* and a = 7.6. Thus t = at, = 15.2 s. If, instead, r„ = 4 x 10"^ s, then / = 19.8 s for recovery to 0. 1 ppm, an increase of only 4.6 s for this tenfold

increase in off time. This interesting result (a simple con- sequence of the fact that In 10 = 2.3) shows the relative insensitivity of the recovery time to changes in the relay

OFF time whenever t„ « f,.. 123
306 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT. VOL. IM-36, NO. 2, JUNE 1987

References

1

1 ] F.J. Wilkins, "Theoretical analysis of the ac/dc transfer difference of

the NPL multijunction thermal converter over the frequency range dc

to 100 kHz, " IEEE Trans. Instrum. Meas. , vol. IM-2I, pp. 334-340,

Nov. 1972.

12] F. L. Hermach and D. R. Flach, "An investigation of multijunction thermal converters," IEEE Trans. Instrum. Meas., vol. IM-25, pp. 524-528, Dec. 1976.

[3] F. L. Hermach, "An investigation of the uncertainties of the NBS ther- mal voltage and current converters," Nat. Bur. Stand. (US), Rep. NBSIR 84-2903, Apr. 1985.

[4] B. D. Inglis, "Errors in ac-dc transfer arising from a dc reversal difference," Metrologia, vol. 17, pp. 111-117, Nov. 1981.

[5] F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency voltage measurements of high accuracy," IEEE Trans. In- strum. Meas., vol. IM-15, pp. 260-268, Dec. 1966.

[6] D. Zhang and Z. Zhang, "Method for reduction of ac-dc transfer error caused by the Thomson effect for the multijunction thermal converter," IEEE Trans. Instrum. Meas., vol. IM-29, pp. 412-415, Dec.

1980.

[7] F. L. Hermach, "Thermal converters as ac-dc transfer standards for current and voltage measurements at audio frequencies," J. Res. Nat. Bur. Stand., vol. 48, pp. 121-138, Feb. 1952.

[8] E. S. Williams, "Thermal voltage converters and comparator for very

accurate ac voltage measurements," J. Res. Nat. Bur. Stand., vol. 75C, pp. 145-154, July-Dec. 1971.

124
524 Section 6

An Investigation of Multijunction Thermal Converters

FRANCIS L. HERMACH, fellow, ieee, and DONALD R. FLACH

Abstract—The relative ac-de differences of a group of multi- junction thermal converters (MJTC's) have been determined over

the frequency range 30 Hz-10 kHz. These MJTC's are of different ranges and were obtained from several sources. Differences were

observed at low frequencies when converters of various ranges were intercompared. For voltage measurements, the use of matched

resistors in series with the MJTC heater resistors greatly reduced

these errors and contributed to the reduction of other errors as

well. It is believed that the average ac-dc difference of this group

is less than 0.3 ppm at 160 Hz and 0.5 ppm up to 10 kHz.

Introduction

MEASUREMENTS of rms current and voltage have been based on a technique which converts

electrical energy into heat and compares this with the heat

produced by a dc signal. Single-junction thermal converters, each consisting of a wire heated by an electric current with a thermocouple to sense the temperature-rise,

have been in use for almost 3 decades at NBS for accurate rms ac-dc transfer measurements. Recently, a potentially more accurate form of the resistor-thermocouple combination, the multijunction thermal converter (MJTC), was

developed at NPL by Wilkins [1]. Verification of these

MJTC's, to 0.5 ppm at 9 V and 1 59 Hz, was initially needed

at NBS for an absolute volt experiment, now well underway. Later, a goal of 1 ppm from 30 Hz to 10 kHz was es- tablished as a marked and useful advance over the present

10 ppm.

Each MJTC has a bifilar heater wire with from 50 to 200

series-connected thermocouples to sense the heater current, overcoming fundamental limitations in single-junction thermal converters [2]. Since actual converters may

depart from the ideal values, experimental comparisons

of converters of different construction, range, etc are desirable to buttress the theoretical accuracy. Unfortunately

there are at present no other suitable kinds of ac-dc

transfer standards to provide additional verification [3]. MJTC's with heaters ranging from 5 to 25 mA and with

rated output EMF's of 30 to 120 mV were obtained from

commercial manufacturers for this investigation, and

50-mA MJTC's were donated by NPL. Techniques were

developed for intercomparing MJTC's with high precision,

and a number of ac-dc errors were uncovered and over- come, to attain the desired goals.

EMF Comparator

The ac-dc difference is defined as 6 = {Qa — Qd)/Qd

where Qa and Qd are the ac and dc inputs which produce

Manuscript received June 29, 1976. The authors are with the Electricity Division, National Bureau of Standards, Washington, DC 20234.

the same output of a thermal converter. A modified Williams EMF comparator [4] was used with a commercial

chopper-type nanovoltmeter to determine A = 6i - 62, for two MJTC's whose heaters were connected either in series as thermal current converters (TCC's) or in parallel as thermal voltage converters (TVC's). A simplified schematic of the measuring circuit is shown in Fig. 1. The ll-kU

three-stage divider D, (adjustable to 1 ppm) and the Lindeck potentiometer P, were first adjusted to bring the

nanovoltmeter N, near zero for both positions of the key

K with dc applied, and were thereafter not disturbed

during an ac-dc determination. S was switched to ac, dc-D

(direct), dc-R (reversed), and AC in succession. In each

case the supply voltage was adjusted to bring N to zero^

in the A position of K and the reading of N in the B position was recorded. As shown in [4], A = 61 — 62 = {Nd —

Na)/n2E2, where Na and Nd are the average voltage

readings with ac and the two directions of dc applied, re- spectively, and n2 is 2 for these square-law MJTC's.

When the dc reversal difference was larger than the

range of N, the reversal compensator (RC) was used. It consisted of an adjustable, O-13-mV dc source, with niegligible change on reversal. It was switched on for one direction of the applied dc current and reversed for the other, so that there was no net change in the average. Carefully timed sequences of ac-dc readings, with 30-s

intervals and less than 10-ms off times were necessary to overcome serious limitations from drifts and fluctuations

in the MJTC's as well as the power supplies. In some cases Johnson noise was 0.3 ppm, and detector noise was 0.6 ppm, peak to peak, in the pass band of the detector. The

MJTC's were placed in a 0.02 m'^ passive oil bath in a temperature controlled room. Even so, some had to be

discarded because of excessive drifts or large switching

transients.

The detector had a 20-dB narrow-band rejection filter centered at 60 Hz. and was sensitive to common-mode

voltage at and near harmonics of its chopper frequency of

94 Hz. Significant ac outputs at twice the input frequency'-^ were observed in MJTC's below 100 Hz (approximately

proportional to 1//'^), and at the input frequency above

1 kHz. Errors from these were reduced by modifying the

filter and connecting a 2-^xF capacitor across the instrument.

To verify the accuracy of the nanovoltmeter, a number

of ac-dc difference measurements were also made with a

' Close settings were not necessary for square law MJTC's. Like the prototype the potentiometer could be connected across £2, but this was

rarely done, except to measure £2-

2 Peak-to-peak ac outputs of up to 0.5 percent, over 1000 times the normally used 0.3-mV range of the detector, were observed at an input

frequency of 30 Hz.

125
HERMACH AND FLACH: MULTIJUNCTION THERMAL CONVERTERS 525

0

Fig. 1. Comparison circuit.

8 PPM

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

-0.5

\ 85rnA

^ lOm

ClOm/

8 2: mA

30 50 100 200 300 500 1000 FREQUENCY IN Hz

Fig. 2. Ac-do differences of TCC's.

photocell galvanometer amplifier as a null detector, in- serting small known voltages in the detector circuit with

the reversal compensator. The results with the two detectors were in agreement to 0.2 ppm or better.

Low-Frequency Errors

Initial tests were made from 30 to 1000 Hz (a range near

5/1 about 160 Hz). Stepupicomparisons as thermal current

converters (TCC's) were made by connecting, for example,

two 5-mA MJTC's in parallel (outputs in series) and con- necting the pair in series with a 10-mA MJTC. The two

5-mA MJTC's were also compared, with their heaters in

series. These tests disclosed alarming differences between

ranges, roughly inversely proportional to frequency, as shown (at rated currents) in Fig. 2, with the 50 mA MJTC's

taken as a base. The cause of these differences is as yet

unknown. The error was affected by shunting the heater

and was opposite in sign for TVC's. Thus it was caused by

a change in the effective resistance of the heater at low

frequencies.

As the frequency of the applied current decreases, the

heater temperature of a thermal converter begins to follow

the double-frequency variation of the instantaneous input

power. This can cause low-frequency errors if the resis- tance is dependent on the temperature. However, the

MJTC heaters are quaternary alloys of mainly NiCr, which

have low temperature coefficients (0 ± 20 ppm/°C). One

pair of MJTC's of the same range had nearly equal lowfrequency errors but opposite temperature coefficients.

The error was generally opposite in sign from that deduced

mathematically by treating all of the heater as a mass at uniform temperature, with all of the heat lost by temperature-independent lateral cooling. The thermal situation

must be much more complex than this, as evidenced by the

measured 1//'^ variation of ac output with heater fre- quency, instead of 1// as predicted by this and other simple

models when the ratio of ac to dc outputs is small.

A prototype MJTC with a NiCr heater from a third

manufacturer had much larger ac dc differences; about

-100 ppm as a TCC and +120 ppm as a TVC at 30 Hz.

This shows rather strikingly the necessity for experimentally verifying new designs. These errors agreed with the

sign predicted by the model, but varied roughly as 1/f in- stead of 1//2 as predicted for low ac outputs.

Ac bridge measurements showed that the low-frequency

insulation resistance between the halves of a heater that

had been cut in the middle was too high to account for the

low-frequency error. Thus the low-frequency error is apparently not caused by the temperature coefficient of re- sistance of the heater or dielectric loss between the halves

of the heater.

A well-known theorem states that the power in a resistor Ri fed by a constant voltage source having a fixed internal

resistance Ri is a maximum when R\ = /?2- From this, as shown in the appendix, the power in the heater of an MJTC used to measure voltage would to the first order be

independent of small changes in the heater resistance if a matched (equal) fixed resistor were connected in series with the heater. Analogously, for measuring current the

power would be independent of small changes with a matched resistor across the heater. Since the thermocouples respond to the power dissipated in the heater, these

interesting applications of the maximum-power-transfer

theorem make possible compensated MJTC's, in which the

first-order errors from changes in effective resistance are eliminated, at the cost of doubling the current and voltage

ranges. Calculations and tests showed that matching to 10

percent is adequate for the desired accuracy. Other Errors

Other errors in voltage measurements arise from Peltier

effects at the junctions between the heater and its lead-in

wires, and skin effect in magnetic leads used in some MJTC's [1], [2]. For a given construction the errors from

Peltier effects are independent of frequency (except at low

frequencies) and current level, and are inversely proportional to the resistance, so that they could in principle be

126
526 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASI IREMKN'i', UECEMHER 1976

TABLE I TVC Comparisons

MJTC MJTC

4 - « 1 -- «2 (ppm) n V 30 Hz 160 Hz 1 kHz 10 kHz B10*2

# B5 2 9 +0 4 -0.2 -0.2 -0.4 B10*44 B5*2 9 +0 6 -0.1 0.0 -0.2

B10*2 810*44 6 +0 1 - 0.0 +0.1

9 0 0 0.0 -0.2 +0.2

C10*2 6 +0 4 +0.

1

+0.1 +0.2

C10*l 6 +0 4 +0.1 +0.1 +0.6

C10**2 BIO 45 6 +0 4 -0 .

1

-0.3 -0 .

1

cio*i 6 +0. 3 0.0 +0. 3 +0.2

B10*2 6 +0 1 +0.2 0.0 +0.4

A50*14 6 +0. 4 0.0 -0.2 A50*15 8 +0.2 o.p -0.1 A50*15 B10*2 9 0. 0 -0.1 +0.2 -

NOTE: B10*2 E ignlf ies a 10 mA MJTC, s erial number 2, of manufact urer B.

B5*2 and B10*2 were mounted in unevacuated glass tubes at NBS.

A50*14 and *15 were in evacuated gla ss tubes w 1th magnet Ic leads

.

All others were in evacuatec copper containers

evaluated by making TVC tests with different series resistors. However the low-frequency variation can be

complicated, and the evaluation is difficult in the presence

of other errors. Instead these and skin-effect errors were

eliminated by specifying MJTC's with copper leads, which

have low thermal emfs against the quaternary heater al- loys.

At higher frequencies (up to 10 kHz), tests disclosed

errors in some of the MJTC's from dielectric losses between the heater and the thermocouples."^ For example,

differences of up to several ppm at 10 kHz could be obtained by changing the ground from one line to the other

in tests of two 5-mA TCC's in series. Dielectric errors were

also greatly reduced by the matched resistors. (See Appendix.)

TVC Tests

Since the needs were for voltage measurements, a group

of 8 MJTC's of 4 different constructions and 3 current

ranges, and with copper leads, were compensated with

matched series resistors of quaternary alloys wound on thin

mica cards.'* Each MJTC with its resistor was mounted in an aluminum box (through which oil could circulate), with

input binding posts and a 2-pin output connector of low

thermal EMF's. They were compared as TVC's at 6 to 9 V,

at 30, 160, 1000, and 10 000 Hz. The results are shown in Table I. At the important frequency of 160 Hz the results are shown more vividly in Fig. 3. The circles identify the

MJTC's and the number adjacent to each arrow gives bh — bt in ppm, where bh and bt are, respectively, the ac-dc

Dielectric losses in the oil were negligible. Two MJTC's, with 50-mA ranges, have magnetic leads and were not used above 1 kHz. A series resistor of 200 Q was used with each, to give

a 10-V range. Tests with other resistors showed that resistance-dependent

low-frequency errors of these MJTC's were less than 0.5 ppm. Series

resistors for the other MJTC's were within 10 percent of the heater re- sistances.

TABLE 11 Assigned AC~DC Differences

6 (ppm) MJTC 30 Hz 16*^ Hz 1 kHz 10 kHz BlO A A 0 .

0

-0 1 -0 2 -0. 2 BIO 45 0 .

0

+0 1 -0 1 0.0

CIO**! +0.3 +0 1 0 0 +0. 3 010*^2 +0.4 0 0 -0 2 0.0

B10**2 0.0 -0 1 -0 1 -0.2

B5 2 -0.5 +0 1 0 0 +0.

1

0.0 0 0 +0 2 A50^15 -0.1 -0 1 +0 2

Fig. 3. TVC's at 160 Hz.

differences of the MJTC's at the head and tail of the arrow.

In the absence of systematic errors the arrows around the

loops should sum to zero within the limits imposed by the

random errors of the comparisons. Alternatively failure to close properly is an indication of systematic errors. Techniques described by Youden [5] can be used to eval- uate networks such as this. The values of b assigned in this way to each MJTC at 160

Hz are shown in each circle, on the assumption that the

average of all is zero. Each value is less than 0.2 ppm, and

the closure errors are very small.

Similar analyses were carried out for the measurements

at other frequencies. The assigned ac-dc differences are shown in the columns marked b^ in Table II. The dis- crepancies are somewhat larger than at 160 Hz but there

is no evidence of systematic error, or of significant residual

low-frequency errors in these compensated MJTC's.

The standard error s (standard deviation of the average

of 4 determinations) in a comparison of 2 MJTC's was 0.12 ppm. Because of the network of comparisons the standard

deviation of an MJTC with respect to the average should

be somewhat less than this.

Series Resistors

The high-quality wire-wound series resistors used with

these MJTC's have excellent ac-dc characteristics at audio

frequencies. The phase angles are small and cause only a

127
HERMACH AND FLACH: MULTIJUNCTION THERMAL CONVERTERS 527

Fig. 4. Reversing bridge.

second-order effect on impedance magnitude. However

there is a possibility of dielectric loss in the insulation at the higher audio frequencies. (The maximum powertransfer theorem does not help here.) To determine this,

the 1/1 inductive-voltage-divider bridge shown in Fig. 4 was set up to compare two nearly equal resistors /2i and R2.

A balance was made for each switch position of this "re- versing bridge" by adjusting m and n with the detector lead

d, as shown, after adjusting the Wagner arms with d

grounded. Most of the bridge errors were eliminated in this way, leaving as the difference equations, (/?2 ~ Ri)/Ri ~

2(Mq — nt,) and 02 — ai = yinia — rrib) where R and a are

the resistive components and the phase angle, respectively,

of the parallel equivalent circuit for each resistor, 7 =

uC^R, and the subscripts a and b refer to the switch positions shown. These simple equations hold to 0.1 ppm if the

phase angles of C3 and m are each less than 0.1 mrad and

(a2 — ai) is less than 1 mrad.^

A l-kl2 resistor of the same type as those used with the

MJTC's was mounted in a metal box, for four-terminalpair ac-dc measurements on Cutkosky's bridges [6] from

160 to 16 000 Hz. This served as a standard for evaluating

two 1-kQ unmounted resistors. These were paralleled to evaluate a 5-section 500-Q card. The 5 sections were in- tercompared and then used to evaluate the 300, 400, and

500-fi cards for the MJTC's. The 1300-12 card for the 5-mA

MJTC was compared with the 1-kfl card in series with 3

100-Q sections.

An upper limit of 10 kHz was imposed by the effects of

lead inductance and of stray capacitance from the cards

to the grounded aluminum boxes in which they were tested

and used. Below this, changes in the parallel resistances were less than 0.1 ppm for drastic changes in the positions

of the resistors. The cases of the inductive voltage dividers

and the shields of all leads of the reversing bridge were

grounded. C3 was a low-loss 3-terminal air capacitor, and

D was a tuned null detector. All bridge components were

commercially available.

Measurements were made at 0.1, 1, and 10 kHz with 8

V appHed. The results are shown in Table III as dg and a, where 5^ = (Rf - Rp)/Rp and a is the phase angle. Rf and

Rp are the parallel-equivalent-circuit resistances at the

frequency / and the base frequency (0.1 kHz), respectively.

For each resistor, a was closely proportional to /, and so is not shown at 1 kHz in the table.

5 The dividers were rated up to 10 kHz. Only the lower dials were changed on reversal, with only moderate accuracy required.

TABLE III Resistance Measurements

R

s 1 kHz 10 kHz MJTC Q i e

6 e

a 6 r

6 c B10*44 500 -0.1 -0.7 -0.55 -0.5 +0.3

B10*45 +0 .

1

+0 . 3 -0.41 +0 .

1

-fl 1 ~\J . X C10*l 300 0.0 +0.5 -0.60 +0.1 +0.2

CIO 2 0.0 0.0 -0.41 -0.1 +0.1

B10*2 400 0.0 +0.4 -0.47 +0.1 -0.3

B5*2 1300 0.0 0.0 -0.26 0.0 +0.1

NOTE: All values of 6 in ppm, a in mrad. See text

.

The effect of these small errors on the magnitude of the current through the MJTC is approximately 5r = 6^/2 -

3aV8. This was negligible at 1 kHz and, except for one

resistor, even at 10 kHz. The average for all of the resistors was less than 0.1 ppm at each frequency, so that no cor- rections need be applied to Table II. The ac-dc differences of a TVC is 5^ = 6^ + dc [2] where

8c is the ac-dc difference of the thermal converter. Since

and dr were evaluated as indicated above, 8c could be

determined for each MJTC. As shown in Table III, each was 0.3 ppm or less at 10-kHz.

Conclusions

This work reaffirms the value of both experimental and

theoretical investigations to evaluate ac-dc differences in an absolute sense, and to determine how well actual ac-dc

transfer standards meet the theoretical ideals. A number

of sources of ac-dc differences were uncovered, and were

eliminated by changes in design, or were compensated for. The agreement between a group of 8 MJTCs which met

these requirements was excellent. Because of this, because

of the significant differences in construction and ranges

of these MJTC's, and because of our long investigation, we

believe it is very unlikely that the average ac-dc difference

of this group exceeds 0.3 ppm at 160 Hz or 0.5 ppm at 30

and 10 000 Hz. This belief is supported by an analysis

based on estimates of the limits of error from such factors as mismatch of series resistors, Peltier-heat exchanges,

ac-dc differences of resistors, systematic errors in the

comparator, and random errors. For the latter we took as the 95 percent confidence limit 2s/\/n, where n is the

number of MJTC's in the group.

A few measurements over several months indicate a

repeatability of 0.3 ppm or better. The long-term stability

of ac-dc difference is as yet unknown, but there is no rea- son to expect drifts greater than this. Extensions to higher voltages and frequencies seem

feasible. Small ac-dc differences in the effective heater

resistance could be compensated with combined shunt and

series resistors, as indicated in the Appendix. A wide range

of film resistors could be mounted in a geometry that

would allow the reactance to be calculated, so that the

powerful comparison technique of [2] could then be applied.

128
IEEE TRANSACTIONS ON. INSTRUMENTATION AND MEASUREMENT. DECEMBER 1976

Appendix

Compensation for Changes in Resistance

When a resistor mR is in series with MJTC heater, + a), a « 1, the current (with no reactance) is

v; ~

(m + l)R{l+a/im + 1)) ^

a \

im+ l)R\ m+ 1/

neglecting higher order terms in a. The power in the heater

is P = PR{l + a)c^ — (1 + a).

(m+l)2i?\ m+1 /

For a fixed V and m, the power is independent of a if m

= 1 (neglecting higher order terms). Thus since the thermocouples respond to the power dissipated in the heater,

a small difference between the effective dc and ac heater

resistances will cause no first-order ac-dc difference in a TVC with a matched series resistor. Matching to 10 percent will reduce a 2 ppm ac-dc difference (from this cause)

to 0.1 ppm.

Similarly for a TCC with mR in parallel with i?(l + a)

the power in the heater is independent of a if m = 1, so that

there will be no first-order ac-dc difference with a matched

shunt.

This compensation wiU be effective, however, only if the

heater is a two-terminal resistor in the measuring circuit,

and if the thermocouples are insensitive to any change in the distribution of the power. For example, an error would

occur with dielectric leakage from the heater to the thermocouple of an MJTC unless the thermocouple output and

one end of the heater were connected together, and the

thermocouple resistance was much less than the insulation

resistance.

Matching provides only a single compensated voltage

or current range. However, for higher voltage ranges a

similar analysis shows that if a resistor nR is connected

across the heater and mR in series with the combination,

compensation is achieved if n = m/(m — 1). Acknowledgment

The authors are very grateful to F. Wilkins for the NPL

MJTC's, to R. Cutkosky for the ac-dc measurements of the

mounted resistor, and to J. Sutcliffe and I. Malcolm of

Guildline Instruments Ltd. for their extended cooperation. References

[1] F. J. Wilkins, "Theoretical analysis of the ac/dc transfer differences

of the NPL thermal converter over the frequency range dc—100 kHz," IEEE Trans. Instrum. Meas., vol. IM-21, pp. 334-,340,

1972.

[2] F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency voltage measurements of high-accuracy," IEEE Trans. Instrum. Meas., vol. IM-15, pp. 260-268, 1966.

[3| F. L. Hermach, "Ac-dc comparators for audio-frequency current and

voltage measurements," to be published.

[4] E. S. Williams, "Thermal voltage converters and comparators for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand., 75C,

pp. 145-154, July-Dec. 1971.

[5] W. J. Youden, "Measurement agreement comparisons," in Proc. 1962 Standards Lab. Conf., NBS Misc. Publ. 248, pp. 147-152, Aug.

1963.

(6) R. Cutkosky, "New NBS measurements of the absolute Farad and

Ohm," IEEE Trans. Instrum. Meas., vol. IM-23, pp. 305-309,

1974.

129
334 Section 7 Theoretical Analysis of the AC/DC Transfer

Difference of the NPL Multijunction

Thermal Convertor Over the Frequency

Range DC to 100 kHz

F. J. WILKINS

>4&s/rflcr-Measurements of the rms values of alternating currents are frequently made by means of thermal transfer instruments. At low and

midfrequencies Thomson heating can have an important influence on

the ac/dc transfer difference of these devices, whereas at higher fre- quencies the variation with frequency of the effective resistance of their heaters is the most decisive factor. The present paper examines the in- fluence of both Thomson heating and the reactive components of the heater on the performance of the NPL multijunction convertor, and sets out the theoretical analysis that supports the claim that its ac/dc transfer difference does not exceed 1 or 2 ppm up to a frequency of 100 kHz.

Nomenclature

6 Temperature rise of heater above ambient

temperature.

6J Temperature rise of point at which a thermocouple crosses the heater. X Thermal conductivity of heater material. A Cross-sectional area of heater. ax Thomson coefficient of the heater material.

/ Heater current. 21 Length of heater between adjacent thermojunctions.

N Number of thermojunctions on heater. 2L = 21N Total length of heater. P Power loss for unit temperature rise from unit length of heater by convection and radiation.

P21, Total power loss by a radiation and convection from the full length of the heater. Pe Total power conducted through an element.

Ti Resistance of unit length of heater. R 1 Total heater resistance. p Resistivity of heater material.

Fjx, Voltage drop across heater. Vx Voltage at point x along heater.

li Inductance/unit length of heater. C2(p) Capacitance between heater and inner housing.

/?2(P) Parallel loss resistance between heater and in- ner housing.

tan 6 Loss angle between heater and inner housing.

Manuscript received June 9, 1972; revised July 13, 1972. The author is with the Division of Electrical Science, National Physical Laboratory, Teddington, Middlesex, England.

® 1972 IEEE. Reprinted, with permission, from IEEE Transactions on

Instrumentation & Measurement. Vol. IM-21, pp. 334-340, November 1972.

C3(p) Capacitance between one half of heater and

the other half. i?3(P) Parallel loss resistance between one half of

heater and the other half, tan 5 2 Loss angle between one half of heater and

the other half. y2 = f'2 + ]'wc2 Admittance/unit length heater to housing.

y^-f^'^ Admittance/unit length one half heater to the other. / Frequency of alternating current.

CO Radian frequency of alternating current. M Relative permeability of heater material.

I. Introduction

THERMAL transfer methods are in common use for the intercomparison of alternating and direct currents. A

fundamental study of the limitations of the single junction

convertor was first given in 1952 by Hermach [1] and sub- sequently the ac/dc transfer difference of the earliest form of

the NPL multijunction thermal convertor at power and the lower audio frequencies, due almost entirely to the effects of

Thomson heating, was estimated to be much less than 1 ppm

[2] . This determination was based on the solution of the equations describing the heat-flow pattern along the heater and the accuracy claimed seemed more than adequate for all ac measurements. Occasionally, however, as in absolute de- terminations, a more detailed knowledge of ac/dc transfer per- formance is required, and moreover in the intervening years

the design of the NPL convertor has been modified so as to improve its ac/dc characteristics. The present analysis there- fore not only supplements the previous considerations by

taking into account the factors affecting the performance of

the convertor at both lower and higher frequencies but also gives a more accurate appraisal of the influence of Thomson

heating on the latest design, which has an increased number

of thermoelements.

II. Brief Description of Convertor

The element or functional part of the multijunction convertor is a twisted bifilar heater 26 mm long on which are supported 220 pairs of series-cormected thermojunctions hav- , ing common cold junction temperatures. The combined out- i

130
WILKINS: AC/DC TRANSFER DIFFERENCE OF NPL MULTIJUNCTION THERMAL CONVERTOR 335

put from the thermoj unctions thus gives an accurate indication of the mean temperature of the heater with respect to that of

the main structural part of the convertor, and being large may

be measured with ease to a few parts in 10'

.

In the earlier form of the convertor the thermocouples were formed as a helix of triangular section with the heater at one comer of the triangle, the support or return copper lead at another, and a glass rod at the third corner as shown in Fig. 1(a). The later elements are rectangular in section, as il- lustrated in Fig. 2(a), and formed on, and thus attached to, a copper rod insulated with ML enamel. The heaters are twisted biTilar lengths of ML insulated manganin or quaternary

alloys; both alloys have low Peltier coefficients against copper

and low Thomson coefficients. Buffer leads of manganin are interposed between the element and thermocouple leads from

each element and the pins in the glass and alumina lead- through assemblies. These buffer leads are taken close to the

structural members in order to make good thermal contact with them and so to prevent the direct flow of heat between

the pins and the element from influencing the temperature rise of the heater. In the early design the through leads were of

nickel-iron, having a high permeability, whereas in the final arrangement they are of copper. Each form of element is mounted in an inner housing as shown in Figs. 1(b) and 2(b). The glass rod of the first design

is located in a groove cut between the two blocks of copper

and thus serves to anchor the element securely, whereas in the second type the copper rod is placed in a channel Ln the surface of the square-section rod. Each form of mounting is such

that the element is surrounded by a copper shield having the same temperature as that of the thermocouple cold junctions.

Both elements have been included in the preceding descrip- tion for almost all the numerical values that are to be quoted

subsequently apply to the early design whereas it is now

thought most future convertors wUl be made to the second.

However the distributed capacitance of each heater to screen and its inductance do not differ appreciably in the two designs

and thus their ac-dc characteristics should be the same at the higher frequencies.

III. Principle of Operation

The convertor is normally operated to give an output volt- age of 100 mV when the temperature rise of the heater is about 15°C. Little heat is therefore lost by radiation, more

especially as the element is surrounded by an aluminized ra- diation shield and thus almost all the heat developed in the heater wire is conducted to the copper base through the thermocouple wires. The thermocouple wires therefore sense

the coldest parts of the heater and in presenting the sole ther- mal impedance to the heat flow determine its temperature rise. Hence the length of the thermocouple wires, the thermal conductivity, and cross-sectional areas of both the copper and

constantan all combine to determine the temperature rise of

the heater, and the cold junction temperatures and relative resistances of the copper and constantan paths control the

variation of the output voltage/°C rise of the heater.

Copper support and rtturn Icodf

Glass /metal seal

C- soldered cor^ncctions

-Twisted bifilor tntcrmcdiate Icods of 0'2mm diameter manganin

(a)

Twisted bifilar heater

Glass rod

(b)

Fig. 1. (a) Triangular element and connecting leads, (b) Triangular element mounted in housing.

IV. Factors Influencing AC/DC Transfer

Characteristics

A. Midfrequency Response

The lowest frequency in this band is that at which the heater

attains a steady and constant temperature for any constant power input and the resulting thermocouple output voltage is almost free from double frequency ripple. Within the wide

range set by this frequency and that at which the reactive components of the heater affect the current distribution along its length Thomson heating is the main cause for the convertor responding differently to alternating and direct currents. The difference between the two responses may be studied,

and an upper limit assigned to its magnitude, by considering

the heat-flow pattern along a single length of heater wire

bounded by two thermal elements as indicated in Fig. 3. When

thermal equilibrium is established the resultant flow across an element 5x is zero and thus

d'^d dd

XA —r - /aj- P6 + /Vi = 0. dx dx

The solution of this equation [1], when the appropriate

boundary conditions are imposed, shows that the heat con- ducted from the heater through each pair of thermocouple

wires is

cosh 2al - cosh 2nl \ sinh 2al J

and it may be convenient at this stage to restate that P

In = XA XA XA

131
336 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1972

through

(a)

Twisted bifilor heater

(b)

Fig. 2. (a) Rectangular element and connecting leads, (b) Rectangular elements mounted in housing.

^=1

P

^ 2/ i

Fig. 3. Section of heater between two thermocouple wires.

Now is much greater than and as a consequence in the previous consideration a convenient, nevertheless justifi- able, approximation was made by substituting m for a and it was then argued that the ac/dc transfer of the multijunction converter was less than 1 part in 10^. An alternative and more precise simplification is derived if each of the hyperbolic functions in the term a {(cosh 2a/ - cosh 2«/)/sinh 2a/} is expanded in terms of its appropriate

series. Subsequent adjustment, with the retention of all terms up to and including /^ , then reduces it to the form

[cosh 2a/ - cosh 2«/

a i

[ sinh 2a/ + |/W +^m2«2/'*}.

This expression includes one term only, namely ^m^/?^/'', that

is dependent on the Thomson coefficient and thus the additional heat conducted through the elements when the heater is

carrying direct current is

2X^1

The temperature of the hot junctions will then be increased over that when the heater is supphed with alternating current

in the ratio

45 l-i/^m2 + |/^w^+^772^«^/'*

and thus the percent ac/dc transfer difference = 9m'^n^l'^. This is transposed into a more convenient form for evaluation

when the appropriate substitutions are made for m and n. The percent ac/dc transfer difference is then

1 P2lV2lRl(4

7 X'p^TV^

For a 0.005-mm heater (i.e., 50-mA heater in NPL design)

typical values for substitution in the preceding expression

are as foUows:

Pzi = 2 X 10'^ W/°C

aj- = 2 X 10"^ V/°C

= 220

V^i = 0.5 V 1 only half heater

/? = 10 J considered

pX = 1 X 10"^ W/°C (constant most

materials).

The calculated percent ac/dc transfer difference is then

132
WILKINS: AC/DC TRANSFER DIFFERENCE OF NPL MULTIJUNCTION THERMAL CONVERTOR 337

=:10"'\ which is an extremely small value and defines the

original claim of an accuracy of better than 1 in 10"^ with

greater certainty. Moreover this is an upper limit as there is an additional compensation, not allowed for in the calculation, due to the bifilar arrangement of the heater. The two

wires are in close thermal contact with each other and each

will therefore have the same temperature gradients along its length and yet the current flows wUl be in opposite senses. The net Thomson heating across any plane will thus tend to be

zero, so that when allowance is made for this further factor the ac/dc transfer difference previously calculated is made

even smaller.

B. Low Frequency Response

The temperature of a heater supplied with an alternating cur- rent of frequency lower than the midfrequencies of the pre- vious section will vary continuously throughout each cycle

and the corresponding thermocouple output voltage will there- fore include a double frequency component. Nevertheless provided the power input and thermocouple output are di- rectly proportional to each other the true rms value of the

alternating current may be determined by equating the mean

of the varying output voltage to the constant voltage de- veloped by a direct current. Nonlinearities in the relation wUl,

however, result in errors in the rms value and the greater the

nonlinearities and temperature excursions the larger wUl be

the errors. The elements or functional part of the thermal converter is mounted in an evacuated container so that the heater does not

lose any heat by convection and moreover, because its temperature is only a few degrees above ambient, its loss by radia- tion is small and proportional to its temperature rise. Almost

all the heat is therefore conducted through the thermal ele- ments to the heat sink, which is the structural part of the convertor. The temperature rise of the heater for a given power

input is therefore determined by the length and cross- sectional areas of the copper and constantan, and the thermal

conductivities of both copper and constantan. Published fig- ures indicate that the conductivity of constantan increases and that of copper decreases with increasing temperature.

However, when due allowance is made for these variations and for the slight loss of heat by radiation, the temperature

rise of the heater over its normal restricted working range can be shown to be directly proportional to the power input.

The thermal EMF of a copper-constantan thermocouple

may be expressed in the form

E = at + ^t^ (a and /3 both positive)

and thus if there were no further compensating factor the per- formance of a convertor would tend to be oversquare, that is the ratio of voltage output/power input would increase with

increasing power input. The thermal EMF of a thermocouple

system formed by the electrochemical deposition of copper on

discrete lengths of constantan is however also dependent on

the resistance of the copper path and may be shown to be

given by the expression

effective EMF = (copper-constantan ^^^^^ thermal EMF),

where Ri is the resistance of each length of copper and i?2 the resistance of the constantan lying underneath it. The

cross-sectional area of the electrochemically deposited copper

is therefore made such that the decrease of the first term with

rising temperature, due to the large temperature coefficient of

copper, will just compensate for the increase in the second and

the output voltage is then directly proportional to power in- put. Under this condition it is claimed that the low frequency

response of the convertor wUl be equally as accurate as at midfrequencies. Any slight departure from the direct proportionality between output voltage and power input will lead

to the greatest errors when the temperature excursions of the heater are largest and wUl only have a slight effect on the ac/dc

transfer difference when the double frequency ripple is small. The responses previously described are dependent on conditions that must be realized whUe the element is being constructed. Thus precautions are taken to ensure that the copper

is of a specified and uniform thickness, that the pressure inside the housing remains at constant low value, and that the loss of heat by radiation is well defined. These various design

features will be described in greater detail in a further paper.

C. High-Frequency Response

In the determinations of the ac/dc transfer differences of the multijunction convertor at low and midfrequencies both the current paths and also the distribution of the current across all sections of the heater have been assumed to be the same for both direct and alternating currents. At much higher fre- quencies these assumptions are no longer valid for the reactive components of the heater and their associated loss resistances influence the pattern of flow of the alternating current to such an extent that it will deviate considerably from that of

the direct current; moreover because of "skin" and "proximity" effects the alternating current is confined to surface

regions of the heater and not distributed uniformly across it. The present theoretical investigation examines, from first principles, how the reactive components of the heater influence

the power dissipation within it when it is carrying a highfrequency alternating current and also assesses, in terms of

the standard formulas of MaxweD and Butterworth, its change

of resistance with frequency due to the concentration of the current in its surface regions. The resultant of both effects determines the ac/dc transfer difference at high frequencies.

1) Theoretical Analysis of the Effects of the Reactive Components of the Bifilar Heater on the Power Dissipation Within

It: Fig. 4 shows a bifilar heater having a distributed impedance Zi/unit length due to its resistance and inductance, dis- tributed admittance 72 /unit length between the heater and

the inner housing, and distributed admittance V3 /unit length

between one half heater wire and the other. The admittances

in each case are due to the capacitances between the respective

parts of the convertor and their corresponding dielectric losses.

133
338 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, NOVEMBER 1972

, I / unit length

i:

1

vyonit length

Heat link connected to one end of heater

Fig. 4. Bifilar heater showing impedance r i /unit length and distributed admittances^j and js/unit length.

Thus

j^h (effective impedance/unit length of heater)

y2 = ^2+ jwc2 (effective admittance/unit length of

heater to case)

y^- ''3'^ j^Cs (effective admittance/unit length heater

to heater).

The impedance/unit length of the heater is determined solely

by the nature, dimensions, and separation of the heater conductors, whereas the effective admittances between heater and case, and between one half heater and the other are de- pendent on the properties and quantity of the materials lying external to the conductors. Thus the enamel on the wire, the adhesive bonding the elements to the heater and also lying between the two heater wires, and the insulation be- tween the thermal elements and base are all of importance in high-frequency considerations. The rate of change of current and voltage with distance

along heater are as follows:

didx

and

dV

dx

When these equations are expressed with respect to the center of the heater they appear as

di

dy

dV

dy

= y2Vy+y^(Vy-V_y)

= Zliy

and the solution to these equations gives

V2L cosh 6 ^ sinh cosh i L sinhcL

Fji, I b sinh by ^

c cosh cy

^ 2zi [coshfeZ, sinhcZ,

and

= Pi +qi

=Zi(v2 + ,V3) = ('' + Mi) [ri +'3 +/<^(C2 + C3)} = ri{r2 +/-3)- W^/l(C2 +C3) + /Cj{/-l(C2 +C3) + r2/i} = P2 +/?2-

The power dissipation within heater +L

h.l^ dy=F{bcL).

The integration is derived by expanding iy in powers of b

and c and then, neglecting terms of the order b^ and it is shown that

F{bcL) = 2Lp I /o' I { 1 - 2L 2 ( p J + p2 /3) + 21^ /3

[3{2p\ +p,p2 +pll5)-(2q] +qiq2 +^72/5)]}

and if we write

P2 =1^2

then

F(bcL)'=2Lp I (3^1 +P2 - [3(2P? +PiP2 +^1/5)

-{2Q]+QiQ2+QllS)])}

and thus the ac/dc transfer difference expressed as a fraction of the "dc" power dissipation = §{3A +P2 [3{2P\+P,P2+Pl/5)

(221 +Gi 22 +21/5)]}.

When pi and P2 are small the ac/dc transfer difference ap- proximates to ILHpi + 1P2) = 2LH^rir2+r,rj3 + ^co^hc2 + ^co^hcs)

and this formula is adequate for considerations of the NPL

converter up to 100 kHz. However both pi and P2 are fre- quency dependent and thus when the performance of the convertor is required at higher frequency levels the more exact formula is necessary; the values of both terms are also de- termined by the heater resistance. In the case of a typical

10 mA NPL thermal converter pi is approximately 5 X 10"^

at 100 kHz, P2 is 1 X 10"^, and the ac/dc difference is then

1.5 X 10"* as shown in Table I. 2) Determination ofReactive Components ofHeater:

a) Inductance/Unit Length: The inductance of the bifilar heater is small, difficult to measure, and is therefore calculated from the standard formula namely: inductance/unit length =

(Mo/4) (2 \oge D/oq + -y) H/m - D is the distance between

134
WILKINS: AC/DC TRANSFER DIFFERENCE OF NPL MULTIJUNCTION THERMAL CONVERTOR 339

TABLE I

Heater-Base (PaiaUel) Half-Heater-Half-Heater (ParaUel) Frequency

(kHz) C2{P) X

10-'2 tan 6

1

(") C3(P) X

10-12 tan 62 ^3(P)

(")

Calculated AC/DC Transfer Difference 50 mA 35 mA ID mA

1 1.2 0.008 1.6 X 10'° 3.3 0.005 1 X 10'° 10 1.2 0.003 5 X 10' 3.3 0.006 1 X 10' 7 X 10"' 1.5 X 10-8 1.5 X IQ-'' 20 1.2 0.002 3X 10' 3.3 0.007 5 X 10^ 1.5 X

10-8 4 X 10"8 3 X

10-'' 50 1.2 0.002 1 X 10' 3.3 0.008 2 X 10^ 3 X 10-8 8 X 10-8 7 X

10-'' 100 1.2 0.002 5 X 10^ 3.3 0.008 1 X iqs 7 X 10-8 1.5 X IQ-'' 1.5 X 10"^

centers and Oq is the radius of the conductors. A value of D

equal to 2ao increased by 5 percent to allow for the enamel

coatings was assumed in all calculations and thus z = +

/a;(2 X IQ-''), where ri equals the resistance of unit length of

heater. b) Admittances of Heater to Case and of One Half-Heater

to the Other: The admittances >'2 and 73 were both calculated from measurements of the capacitance and loss angle made

directly on elements of triangular section that were mounted

in the normal manner in inner housings but without the added

complication of the outer housing. The capacitance and

tan 5 between the complete heater and the housing were first determined and then the looped end of the heater was cut and

the capacitance and loss angel between the two halves of the heater measured independently. The measurements were re- peated on each of three sizes of heater, at each of a number of

frequencies up to 100 kHz, so that the variation of the ac/dc

difference with frequency of each could be ascertained. The

scatter in capacitance and tan 5 values between one heater and

another was not large enough to affect the calculated ac/dc

difference significantly so that only mean values are quoted

in the following. All measurements gave parallel capacitance

and resistance and these values were first transposed to give the required admittances; the results and conclusions are summarized in Table I and should also apply to the rectangular elements.

3) Calculated AC/DC Transfer Difference Due to Reactive

Components: The heater resistances are 50 mA—15 Q., 35 mA-40 Q., 10 mA-360 Q., and = 2 X 10"^ H/m. The

ac/dc difference is

21' i^r.r^ + \ryr^ + fco^/jCa + WhCi}.

The preceding analysis, therefore, leads to the conclusion

that the ac/dc differences due to the reactive components of

the heater are small even at frequencies as high as 100 kHz.

They are, nevertheless, of greater magnitude than any ac/dc

differences caused by Thomson heating.

4) Determination of Change of Heater Resistance With Fre- quency: The change in resistance of a single heater wire with

frequency, the skin effect, is determined from the standard

Maxwell formula [3] namely

Rdc 12 180

where

^

2v^alfii

10'

The proximity of the second current carrying wire, which

will give rise to an asymmetric current distribution around the

surface of the first wire, causes an additional increase in the

effective resistance of the heater. This further increase is cal- culated from a development of Butterworth's equation by

Arnold [4] in which the fractional eddy current losses in any

conductor are expressed as a product of the skin effect and a factor that is dependent on the form of the conductor

system. For a bifilar unit such as the converter heater 7 is shown to be 2 and thus the increase in resistance due to the

skin effect must be doubled to allow for the proximity effect. Insertion of the known constants of manganin in the pre- ceding formula allowing for the proximity effect, shows that the resistance of a 0.05-mm wire will be 6 parts in 10* higher

at 100 kHz than with direct current, the corresponding

changes for a 0.03-mm diameter wire will be 6 parts in 10',

and for a 0.012-mm connecting wire 2 parts in 10^. The latter, however, is only a small part of the total resistance and may be

neglected.

5j Calculated Overall ACIDC Transfer Difference: The main

source of ac/dc transfer difference at frequencies of the order

of 100 kHz is therefore that resulting from the reactive components, but even in the case of a 10-mA converter and the ac/dc transfer difference probably does not exceed more than 2 parts in 10*.

6) Other Sources of High Frequency ACIDC Transfer Dif- ference That Could Affect Measurement of Voltage by Convenor: In the earlier type of converter the leads through the matched glass seal were of a nickel-iron alloy having a relative permeability of about 10 000. The resistance of the leads was

therefore very frequency dependent and even at 5 kHz the change of resistance from the dc value was sufficient to cause

appreciable errors in some voltage measurements. In the later design the corresponding leads are of copper and the calculated change of their resistance even at 100 kHz is so small that it should not impair the overall accuracy of the converter for any

voltage measurement. The one serious limitation of the earlier form of the NPL converter has thus been rectified. Eddy currents induced in structural components might be

another source of error for in order to define the temperature

of the intermediate leads they are taken in close proximity to

135
340 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-21, NO. 4, NOVEMBER 1972

to the Stout copper pillar on which the element is mounted.

These leads are closely twinned and should not therefore have

any extraneous magnetic field surrounding them to cause in- duced currents. Indeed observations indicate there is probably no ac/dc difference due to this effect; it is nevertheless an issue that could be affected by slight changes in constructional

procedure.

Lead inductance is yet another possible cause of ac/dc

transfer difference at high frequencies. Thus, while all leads

to and from the welded joints are carefully disposed to reduce

loops to the minimum possible, they cannot be eliminated

completely and at some frequency level their inductance will begin to limit the accuracy of the convertor for voltage measurement.

V. Conclusion

The analysis set out in the paper has shown that the ac/dc

transfer difference of the NPL multijunction thermal convertor is almost completely free from the effects of Thomson

heating and does not exceed 2 ppm at 100 kHz due to the change with frequency of the effective heater resistance.

Acknowledgment

The author wishes to acknowledge the help of F. L Hermach

of NBS who suggested the revised approach of Section IV-A,

of G. F. MUler and B. T. Hinde of NPL for their solution to the equations developed in Section IV-C-1 and of G.H. Rayner

of NPL for his capacitance measurements given in Section IV-C-3.

References

[1] F. L. Hermach, "Thermal converters as ac/dc transfer standards

for current and voltage measurements at audio frequencies,"

J. Res. Nat. Bur. Stand., vol 48, p. 121, 1952.

[2] F. J. Wilkins, T. A. Deacon, and R. S. Becker, "Multijunction ther- mal convertor-An accurate dc/ac transfer instrument," Proc. Inst. Elec. Eng. (London), voL 112, p. 4, 1965.

[3] Golding and Widdis, Electrical Measurements and Measuring In- struments. London: Pitman and Sons Ltd., pp. 201-3.

[4] A. H. M. Arnold, "Alternating current resistance standards," /'roc. Inst. Elec. Eng. (London), voL 100, pt. 2, no. 75, 1953.

136
Section 8

TKermal Converters for Audio-Frequency Voltage

Measurements of High Accuracy

F. L. HERMACH, fellow, ieee, and E. S. WILLIAMS

Abstract—The ac-dc differences of a reference group of thermoelements have been evaluated at audio frequencies to a few parts per

million (ppm) at currents from 5 to 20 mA. A technique for comparing

the ac-dc differences of two thermoelements with an luicertainty of about 2 ppm has been developed. Two 5 mA thermoelements are used with a plug-in set of resistors of computable reactances to form

thermal voltage converters for voltage measurements. With this same

technique adjacent ranges of these converters can be compared to step up from O.S to 500 V to better than 10 ppm.

L Introduction

IN

PRINCIPLE, the current balances which have

been developed to determine the ampere in absolute measure with direct current could also be used with

alternating current. However, the most stable electrical standards are still the dc standard cell, and 1 ohm resistors, which are not usually designed for ac use. Thus,

ac measurements of current and voltage have, in practice, been based on rms ac-dc transfer standards or comparators. For many decades these were electrodynamic or electrostatic instruments, designed to have the same torque constant with either alternating or direct current (or voltage) applied, making it easier to obtain

a wider frequency range than with instruments like the

current balance which must have a computable torque

constant. More recently, electrothermic instruments

such as thermocouple transfer standards and bolometer

or thermistor bridges have been used. All of these instruments have been highly developed in the national laboratories of several countries to make ac measurements

at audio frequencies to about 100 ppm in combination

with dc standards. The combination possesses a number

of advantages over ac potentiometers (which must also

include a transfer instrument), and over electronic and

other squaring circuits (which must have exact squarelaw characteristics in two quadrants).

Recent advances in operational amplifiers and inductive voltage dividers have made possible wide-range ac power sources with linearity and day-to-day stability

better than 100 ppm. This implies a need for better

accuracy in basic ac-dc transfer standards.

At this accuracy, conventional forms of electrodynamic transfer instruments are limited to power fre- quencies by the effects of inductive and capacitive reManuscript received June 23, 1966. This paper was presented at

the 1966 Conference on Precision Electromagnetic Measurements,

Boulder, Colo. This work was supported in part by the Army Metrol- ogy and Calibration Center, Frankford Arsenal, and the Metrology

Engineering Center, Bureau of Naval Weapons, Pomona, Calif. The authors are with the Electrical Instruments Section, Electricity Division, National Bureau of Standards, Gaithersburg, Md.

actances in their coils, and electrostatic instruments

have a serious minimum-voltage limitation because of their low torque- weight ratio. Both would require new

means of sensing deflections to obtain precisions of 10

ppm or better. This could be done by measuring the

reactance, which is a function of the deflection of the

instrument. However, a major difficulty would remain —^that of vibrations from external forces or from the

cyclic variations of the ac driving torque. Experience

has shown that it is often difficult to control spurious

torques caused by minor mechanical resonances. It is possible that floating the moving system in a liquid of the same density would minimize these difificulties. However, it has seemed to the authors that improved

thermal converters (thermocouple instruments) offer better promise of obtaining an ac-dc transfer accuracy

of 10 ppm or better, and considerable eff'ort has been

concentrated on them over the past five years. This

paper may be considered as a progress report describing

the present group of reference thermoelements and coaxial thermal voltage converters, and the techniques

and equipment which have been developed to compare

them to a few ppm.

II. Thermal Converters

A thermoelement (TE), defined as the simplest type

of thermal converter [l], consists of a heater and ther- mocouple. In its usual form, the heater is a short,

straight wire suspended between two supporting leadin wires in an evacuated glass bulb. The hot junction

of a thermocouple is fastened to the midpoint of the

heater, and is electrically insulated from it with a small

bead. The thermocouple EMF (about 5 to 10 mV for a conventional TE at rated current)' is then a measure

of the heater current. For voltage measurements, a re- sistor is connected in series with the heater to form a thermal voltage converter [l]. At the National Bureau of Standards, fourteen TEs

form the present reference group for ac-dc difference.

Twelve of them (made by four different manufacturers)

are of conventional design but with heaters of Evanohm

or Karma (modified nickel-chromium alloys), which

have low thermoelectric effects.^ They were chosen from

' The international system (SI) of units with its symbols is used in

this paper. Because of their frequent occurrence, the word thermoele- ment is abbreviated as TE and thermal voltage converter as T\'C in this paper.

^ Evanohm and Karma are registered trademarks of the Wilbur

B. Driver Company and the Driver-Harris Company, respectively. (See Acknowledgments.)

137
HERMACH AND WILLIAMS: THERMAL CONVERTERS FOR AF VOLTAGE MEASUREMENTS 261

larger groups on the basis of low dc reversal differences

and high bead resistances, and are all mounted in one aluminum enclosure lined with foamed plastic for thermal insulation. Each is supported by its heater

leads, which are soldered directly to binding posts

mounted on the low-loss phenolic top plate of the enclosure, and its thermocouple output leads, which are

soldered to two-pin connectors. The other two thermal

current converters are of a radically different design

having many thermocouples (40 or more) attached to a

bifilar (doubled-back) heater [2]. They were obtained

from their inventor, F. J. Wilkins of the National Physical Laboratory in England.

The fourteen TEs are identified in Table I, along

with certain pertinent characteristics, as well as the ac- dc differences assigned to them at audio frequencies as a result of the tests to be described.

TABLE I Reference Thermoelements

Rated Dc Approx. Bead Ad-dc differ- Identi- Curre- heater resis- ence (ppm)

fication rent EMF versal t length tance J mA mV ppm mm GQ 20 Hz 2 kHzll NPL 14* 20 25 <10 >1 0 0 NPL 15* 20 25 <10 >1 0 0 A20E3 20 5 40 5 >1 0 0 A20E6 20 5 60 5 >1 0 -1

A10E18 10 8 130 5 0.3 +6 -Fl A10E19 10 8 160 5 0.5 0 -1

WlOEl 10 7 200 8 0.6 + 1 +3

W10E2 10 7 <10 8 0.8 -2 0 B10E45 10 8 160 4 >1 -8 -2

A5E8 5 7 10 5 0.7 + 1 + 1 A5E9 5 6 40 5 0.6 0 -1

F5K9 5 4 20 5 0.5 -3 -2

F5K13 5 4 260 5 0.5 -2 0 B5E77 5 8 110 6 >1 +6 0

* These can be used at much higher currents.

t A/// for the same EMF.

i At 2 kHz and rated /.

II Same value assigned at 20 kHz.

III. Thermal Voltage Converters (TVCs)

Each of a new set of TVCs developed at NBS for this work consists of one or more cylindrical metal-film re- sistors in a coaxial metal cylinder. The resistors can

be connected in series with one of two 5 mA TEs

mounted in separate cylinders so that the output EMF

is then a measure of the input voltage. Like an earlier set developed at NBS [3], the effect of the reactance on the frequency response can be computed at least roughly from the simple geometry. Adjacent ranges of these TVCs may be compared with greater certainty

than those of the older design, which had integrally

mounted TEs.

As shown in Fig. 1, a single two-watt resistor is used

for each range below 200 V. The higher-voltage resistor

units have two or more resistors in larger cylinders, with

two adjustable inner shields to control capacitance currents, as shown in Fig. 1, and have ports and baffles to permit the controlled flow of cooling air from the laboratory compressed-air supply.

Pertinent data on the TVCs, and the ac-dc differences

assigned to each range are given in Table II. The TVCs

are shown in Fig. 2.

IV. EMF Comparator

The ac-dc differences of a transfer instrument may be

defined as

where Xa and Xd are the magnitudes of the ac and dc

quantities that are required to give the same response

(such as deflection or output EMF) of the instrument.

(Normally, the average for the two directions of the dc i quantity gives the best measure of the dc reference.)

This is a useful definition, for the ac quantity for a

j given response is then simply = Xd(l -fS). As is well i known, the instrument may be used as an ac-dc transfer

standard simply by observing the response with the ac quantity applied, then measuring (with external stan- dards) the average for the two dc quantities required to obtain the same response, thus avoiding many of the

limitations of an ordinary instrument.

At the National Bureau of Standards, TEs are evaluated in pairs with the EMF comparator shown in Fig. 3, to determine the differences in their ac-dc transfer

performance, i.e., di — 8z as defined previously. TVCs are connected in parallel and evaluated in the same way.

With the desired current or voltage applied, R\ and R2

are adjusted in a preliminary balance to obtain a near

j null on the detector Di. Then, in regular succession, : alternating-, direct-, reversed direct-, and alternating- ' currents are applied to the TEs at nearly equal time

intervals and without changing Ri or R2. In each case, the current is adjusted to obtain the same output EMF,

E\ of TEi, as indicated by an auxiliary Lindeck potentiometer (P with its detector D2), and the resulting deflections of Di.are observed. Then, as shown in Appendix

I, the relative ac-dc difference is 81-82 = s(Da - Dd)

where Da and Dd are the averages of Di with alternating

and direct currents applied, respectively. The sensitivity factor 5 can be determined in one of several ways, as indicated in the Appendix, such as by changing the cur- rent in the heater of TE2 by a small known amount and

observing the resulting deflection of Di. , The sequence of readings, nearly equally spaced in time, greatly reduces errors from drifts, temperature

changes, etc., in either TE and from either constant or slowly changing thermal EMFs in the comparator. The f comparator accommodates any reasonable range of r thermocouple EMFs, which need not be equal, and the

small currents (up to 20 juA) in the thermocouple circuit

j do not affect the measured values.

With this circuit, small fluctuations in the supply

j (heater) current do not appreciably change the deflec- tion if the two TEs have reasonably equal proportional | changes in EMF (AE/E), with a given change in heater : current and reasonably equal time constants. Although
262 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT DECEMBER

Fig. 1. Essentials of thermal voltage converters (construction details not shown).

A Thermoelement unit, with 5 mA 100 Q TE, for 0.5 V, in brass

cylinder, 4 cm long and 5.1 cm diameter, with brass end

plate D. B Low-voltage resistor unit (1 to 100 V) with one 2-watt metal

film resistor R in brass cylinder 11 cm long and 5.1 cm di- ameter. C High-voltage resistor unit (200-500 V) with multiple 2-watt

metal-film resistors, in brass cylinder 18 cm long and 6.4 cm

diameter with brass end plate D. Air inlet at I, outlet holes at other end (not shown).

Si, 82 Inner brass shields 5.5 cm long, 5.1 cm diameter, connected

as shown, with no end plates. can be moved parallel to

axis and locked in place. P Non-conducting air-baffle plate with holes. E Coaxial connector GR874.

F Two-pin output connector AN lOSL.

it would be difficult to calculate the resultant "stabilization factor" of the circuit, in practice, fluctuations are much less than those observed when the output EMF

of each TE is balanced against a constant voltage

source, as in earlier comparators [4]. Thus, with this circuit and technique, TEs may be intercompared with an imprecision which is much less than the fluctuations

of the supplies used.

Ri consists of ten 100 Q resistors mounted on an enclosed-contact silver-alloy switch whose thermal EMFs

are much less than 0.1 ^iV when the wiper is stationary.

R2 is a 100 fi, 10-turn helical resistor of manganin,

with a manganin wiper to minimize thermal EAiFs.

Detector Di consists of a commercially available primary galvanometer, incorporating photocells in an adjustable negative-feedback circuit, which serves as a

current amplifier to a secondary galvanometer. The

scale factor of the combination is about 10 pA/mm,

and the input resistance (which is dependent on the

negative feedback) ranges from 200 to 3000 12. The

primary galvanometer is liquid filled to reduce disturbances from mechanical shocks and vibrations and, with

its period of two seconds, is an excellent mechanical lowpass filter. A similar pair of galvanometers is used for D2.

Both secondary light beams appear on one scale so that

the observer may quickly read the deflection of Di if it

is stationary and if D2 is in the desired range. (Residual

TABLE II NBS "Model F" Thermal Voltage Converters

Range V

Resistors

i otai

k\1\

hr (ppm)t

IN 0. (each) 2 kHz 20 kHz 50 kHz

D A* none U . 1 + 1 -1-1 + 15 n\J . D-D ^ R* none \) . I -4 -2 + 11

1 0.1 n 0 U . z -2 0 + 11

9 1 0.3 \) .4 -3 — 2 + 3

1,

} 0.5 0 .6 - 1 0 + 3

c 0.9 1 .0 — I 0 _ 1 1 .9 L .0 0 + 1 - 1 20 3.9 4.0 0 + 1 - 1 oU 6 6.1 + 1 + 1 - 2 50 10 10 + 1 0 - 4 100 20 20 + 1 0 - 7 200 4 40 40 + 1 0 - 7 300 6 90 60 + 1 0 - 3 500 8 200 100 + 1 -6 + 13

* TEs A and B as TVCs, without series resistors,

t Including 0.5 V TE. Resistors connected in series-parallel for higher ranges.

X 6r is the impedance error only (see text) computed from Table

III with either TP. of pair #1. A similar table with the #2 pair differs from it by not more than 1 ppm at 2 and 20 kHz except at 0.5 V

range.

Fig. 2. Thermal voltage converters.

I

Fig. 3. Essentials of EMF comparator.

139
1966 HERMACH AND WILLIAMS: THERMAL CONVERTERS FOR AF VOLTAGE MEASUREMENTS 263

deflections of are less critical if the TEs are well matched.)

All of the components of the comparator except the

detectors are mounted in a metal box behind a metal

panel to provide electrostatic shielding, and the box is lined with foamed plastic to reduce temperature fluctuations. The metal cases of the primary galvanometers are connected to this box by the shields of two-conductor

cables. With Ri and Ri at their midpoints and each of the

thermocouple inputs to the EMF comparator shortcircuited, peak-to-peak fluctuations in the secondary

galvanometer of D\ in periods of 10 seconds amount to 1 mm at the gain setting normally used. With 7 mV at each thermocouple input, this corresponds to a value of

5i — §2 of 2 ppm. This is only about four times the calculated Johnson-noise value for this circuit. A comparison of two TEs (or TVCs) usually consists

of four or more determinations of 5i — 62 by the method

described. Three times the standard error (standard deviation of the average) computed from a number of typical series of such measurements is about 2 ppm.

This is a reasonable measure of the uncertainty in the

comparator. Comparisons of various pairs taken from

three TEs, with the TEs interchanged in the comparator

and with the EMF of first one TE and then the other

held constant in each sequence, have disclosed no systematic error as large as 1 ppm.' These measurements were made with commercially available dc and ac voltage

sources, having peak-to-peak instabilities, observed with

a TVC in 10-second intervals, of about 10 and 20 ppm,

respectively, at 2 kHz.

A TVC can be used for ac measurements with appropriate dc standards by balancing its output EMF

against an adjustable stable voltage source such as the

potentiometer P. To reduce errors from drifts, a similar

sequence of ac and dc readings can be used, with the

setting of P fixed, and the direct voltage Vd preset and

accurately known. Then

Fa = Fd[l + 5 + s{Da - Dd)]

where s, the sensitivity factor, is determined as before.

V. Comparisons of TEs

The fourteen reference TEs were compared, using

sinusoidal currents at frequencies of 20 Hz, and 2 and

20 kHz. All of these measurements (as well as the comparisons of TVCs) were made in a shielded room to reduce electromagnetic interference, which is particularly troublesome with TEs because of their high sensi- tivity and very wide frequency range. The step down

from one current to the next was made by connecting

' If the net dc reversal difference of the two TEs is more than 300 ppm, an error is introduced by nonlinearities in D\. This can be mini- mized by taking determinations with first one and then the other TE

EMF held constant.

the heaters of two low-current TEs, a and b, in parallel

and their thermocouples in series aiding, and considering the combination as a single TE connected in the TEi

position of Fig. 3.^ As shown in Appendix II, if the

TEs are reasonably well matched, the ac-dc difiference of the parallel combination is simply 5p=(6a + 56)/2

to better than 1 ppm. From the test, bp — b2 = Ki where

hi is the ac-dc difTerence of the high range TE in position

2, and K\ = s{Da — Dd). The two low-current TEs are then connected in series and compared to determine

Ki = h^-hb. Thus, 5a = i^i +52+if2/2 and h = Ki^h2

-Kill.

The network of comparisons at 2 kHz and the results are shown in Fig. 4. Here the circles identify individual

TEs and the rectangles represent a pair in parallel. The

arrows indicate comparisons, and the numbers on them

are the values of the relative ac-dc difTerences in ppm;

i.e., the ac-dc difference of the TE at the point of the arrow is greater or less than that at the tail by the

amount shown. These tests have extended over a period

of three years during the development of this project;

in many cases, each number is the average of two or more tests, each consisting of four or more sequences

of ac-dc readings already described.^

As indicated by Wilkins, the ac-dc difiference of the

two NPL TEs should, theoretically, be less than 1 ppm.

In tests at NBS at 20 mA, the two agreed to better than

0.2 ppm with a 3cr imprecision of 0.3 ppm.^ Thus, it is reasonable as well as convenient to start with an assignment of zero ppm to the ac-dc differences of each. Then,

following principles given by Youden [5] in weighting

the various paths in a network, it is possible to calculate

the ac-dc dififerences of the other TEs. The resulting

values are given to the nearest ppm in each circle and

in Table I. The average for all of the conventional TEs is only — 0.3 ppm. Thus, these simple TEs agree with the more

complicated multij unction converters very closely, and

essentially the same values would have been obtained

for each TE on the alternative basis of assigning zero to the average of the entire group.

A complete but less redundant chain of measure- ments was carried out at 20 Hz to evaluate low-frequency errors of each TE. The precision was less at that

frequency because of much larger instability in the ac

source. The results of the values assigned on the basis

^ of zero error for the NPL TEs are given in Table I. \ Not enough TEs were intercompared at 20 kHz to assign independent values to each, because ac-dc dif-

* To minimize certain aberrations discussed in Section IX, resisj tors of at least five times the heater resistance were added to each of the heaters before paralleling them. '

' Because measurements were made over an extended time in a ( developing program, the network is not symmetrical.

' The EMF was higher and a higher detector sensitivity could be

used than with other TEs since the two TEs were well matched, so that an extraordinarily high precision was attained.

140
264 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT DECEMBER

Fig. 4. Intercomparisons of NBS TEs, and assigned ac-dc

differences in ppm at 2 kHz.

ferences from all known sources [3] should be the same

as at 2 kHz within 1 ppm. No measured values at the

two frequencies differed by more than 2 ppm. Thus, the same values were assigned to each TE at the higher fre- quency.

VI. Comparisons of TVCs

The TVCs were designed so that adjacent ranges

could be compared to step up and down the voltage

scale. Their nearly identical 5 mA TEs were each first compared with one of the reference group of TEs as current converters, i.e., with the heaters connected in

series. They were then paralleled through a " T" connector

and compared with each other as voltage converters

(basic range 0.5 V). The lowest resistor (see Table 2) was then connected in series with one TE, say TE^,

to make a one-volt TVC. This was connected in parallel

with the other, TEs, and compared at 0.5 volt. A twovolt TVC was then formed with the next resistor and

TEfc, and compared with the one-volt TVC at 1 V. This

process was continued until the highest range was

reached. If the ac-dc difference of each combination is independent of the voltage level, the ac-dc difference of any range may be found in terms of one as a reference.

This requirement is easily met if the TEs have no signifi- cant errors, for the reactance errors of each resistor unit

should be independent of voltage. The validity of this assumption itself can easily be verified by making each

comparison at two voltage levels.

If we define Zt = Vi/Io for a TVC, where F,- is the

magnitude of the applied voltage and h is the magnitude

of the current through the TE, then, from the definition

for ac-dc difference, applied to a TVC,

Via — Vid loaZt — lodRl

5„ = =

Vid lodRi

where the subscripts a and d refer to alternating and

direct current (or voltage). Since loa and hd produce

equal output EMFs, /<,„ = /(,d(l + 6c). If we let Zt = i?,(l-f 5r), we get, if second-order terms are negligible,

5„ = Sr + 5<;. Thus, bv can be determined if 5^ can be deduced from the comparisons of adjacent ranges, since

5c can be evaluated in terms of the reference group of TEs.

Two complete sets of adjacent-range comparisons

from 0.5 to 500 V were carried out at frequencies of 2, 20, and 50 kHz, with two different pairs of 5 mA TEs.

These TEs had 80Ni20Cr heaters but were otherwise

similar to those in the reference group. Measured values

of 5 as current converters were — 1 or — 2 ppm, except at 20 Hz where they ranged from —6 to —9 ppm.^ The

results of the adjacent-range tests given in Table III showed excellent precision. In addition, the inter-range

differences were less than 2 ppm through 20 kHz, except

for the lowest and highest ranges. The differences of a few ppm observed on the lowest ranges were rather

puzzling and will be discussed in Section IX. They

were essentially independent of frequency through 20

kHz but were negligible above the 1 volt range.

TABLE III Comparisons of Thermal Voltage Converters

TVC 1 TVC 2 VoltaV

5i — 62 in ppm

2 kHz 20 kHz 50 kHz

#1* m #1 #2 #1 #2 A B 0 5 +2

B A 0. 5 -3 -3 -4 -3 -4 -4

1-B A 0. 5 +2 0 -2

1-A B 0. 5 +2 +2 + 1 0 +3

2-B 1-A 1 -1 -2 -1 -8 -4

3-A 2-B 2 +2 +2 -fl 0 -1

5-B 3-A 3 0 0 0 -4 0

10-

A

5-B 5 + \ -fl -f-1 0 0 -2

20-B 10-

A

10 0 0 + 1 0 -1

30-A 20-B 20 -1-1 0 0 -1 0 50-B 30-A 30 0 -1 0 -2 0

100-

A

50-B 50 0 0 0 -3 -2

200-B 100-

A

100 0 -f 1 0 -1 0 -F4 300-A 200-B 200 0 0 0 -F4 +6

500-B 300-A 300 0 -6 -5 + 16 -F8

* Test #1—With one pair of 5 mA, 100 ohm TEs.

t Test #2—With another pair of 5 mA, 100 ohm TEs. These tests were made three months after those with pair #1.

Except for this, the results of the comparisons are in excellent agreement with computed values (see Section

VIII), which show that 5r should be less than 1 ppm

through the 100 V range. It is extremely unlikely that

any single source or combination of sources of ac-dc

difference should produce the same 5r in each of these

ranges. Thus, the authors believe that, from the con- cordance of theory and experiment, it is safe to assign zero ppm to the average of the values of 5^ from the 2 V

' These TEs, which were the only ones available at the time with

the correct heater resistance, will be replaced by others.

141
1966 HERMACH AND WILLIAMS: THERMAL CONVERTERS FOR AF VOLTAGE MEASUREMENTS 265

through the 100 V ranges in establishing the values given

in Table II from the comparisons of Table III. The comparisons at 50 kHz (which were less precise) were made to verify that ac-dc differences were not

sharply dependent on frequency at 20 kHz. They indi- cate that excellent accuracy should be expected for most

ranges at even higher frequencies.

A complete set of measurements at 75 percent of rated voltage was also made at 2 kHz with TE pair

#1 in order to investigate voltage coefficients. These

results agreed with the rated-voltage measurements to

1 ppm, with no systematic difference.

A completely independent verification of the accuracy

of these reference TVCs has come from some careful

comparisons by Flach and Marzetta [8] of a "time-gate"

peak ac-to-dc voltage comparator (designed by Marzetta) with an rms comparator which incorporated a TVC that had been calibrated with the reference set. In a long series of tests at 400 and 1000 Hz, where the

error in the comparison introduced by distortion of the ac source was less than 10 ppm, the two comparators

agreed on the average to 4 and 8 ppm, respectively, at the two frequencies.

VII. Thermoelectric Errors

Hermach's treatment of the ac-dc errors arising from

Thomson effects in the heater of a conventional TE [4]

has been extended by Widdis [6] to include the effects

of Peltier heating, and to correct for the heat abstracted

by the thermocouple. Widdis' expression for the Thomson ac-dc difference, bt = oi'^Q/^2pk, is the same as that

found by Hermach for high-range TEs, where a., p, and

k are, respectively, the Thomson coefficient, electrical

resistivity, and thermal conductivity of the heater

alloy, and 6 is the midpoint temperature rise (100°C to 150°C in a conventional TE). The error can thus be

computed reasonably well if a can be determined. The

Thomson heating of copper is reasonably well known

[7], and is not much affected by moderate temperature

elevations or by the cold working of ordinary handling.

Thus, this metal can serve as a practical reference. The

authors have calculated the Seebeck and the absolute

Thomson coefficients of a number of pertinent alloys

from careful measurements of the EMF vs. temperature

of thermocouples formed from each of these alloys and

copper. The results are shown in Table IV, along with

the calculated ac-dc difference 8t (using Widdis' for- mula) in a TE having midpoint heater temperature rise of 150°C (giving a 10 mV output with a typical NiCrconstantan thermocouple). With Evanohm or Karma,

8t should not exceed 1 ppm. At an output of 5 mV it should be less than 1 ppm, since both d and a are

smaller.

Additional measurements indicated that the thermoelectric coefficients may be sensitive to cold working

and may be changed by exposure to high temperature.

For Evanohm and Karma significant changes were obTABLE IV Thermoelectric Data and Errors

Calculated ErrorinaTE||

Metal

or Alloy

a, ii\ / yJ^ at 4UU K. o MV/ C

Dc

reversal** Ac-dc

difference A//7, ppm &i, ppm

Copper -2.0

Evanohm -1.5 +0.3t 200 -1

Karma -1.0 150 <1

Manganin -3. Of -1.8t 400 -4

80Ni20Cr -3.9 +4.4 500 -7

60Nil6Cr24Fe -2.3 -fO.8 300 -2

* a (Thomson coefficient) is proportional to the absolute temperature.

t 5 is Seebeck coefficient vs. copper.

j This was quite dependent on heat treatment.

II From Thomson heating, with 0=15O°C and pk = 25 X

10~* VV°C at/00°K, the weighted average temperature of the heater.

** With the bead off center (see text).

served at 800°C, but not at 400°C. Thus, care must be

taken in controlling the temperature of the wire near welded joints and when out-gassing during the construction of TEs. Here the use of TEs in the reference group

by four different manufacturers having differing techniques provided powerful assurance against unknown

errors. As indicated by Wilkins, the thermoelectric errors are greatly reduced in the NPL TEs because of the bifilar heater and the low temperature rise (about 10°C at 20

mA in the present TEs). The resulting error should be

much less than 1 ppm, even with nickel-chromium

heaters.

Rather unfortunately, commercial TEs have lead-in

wires (see Fig. 1), through the glass bulb to the heater,

of a copper-coated nickel alloy to match the thermal

expansion coefficient of the glass. The combination has

a rather uncertain but fairly large Seebeck coefficient

vs. the heater alloys used. With direct current applied,

Peltier heating and cooling of the junctions of these leadin wires and the heater can cause a temperature differ- ence (+ at one junction and — at the other). Most of this heat is conducted by the lead-in to the thick glass press or disk which forms the base of the glass bulb. This can

be considered as a "heat sink" so that the first-order

junction temperature change is Ad = TIS/K, where T

is the absolute temperature, / is the current, 5 is the

Seebeck coefficient, and K is the thermal conductance

of the lead-in. Widdis showed that this leads to an ac-dc

difference term 8tp = aAd/ Vh where a is the Thomson

coefficient and Vh is the heater voltage. There may, in addition, be a second-order term at the midpoint be- cause the Peltier heat itself depends on the junction

temperature. This can be calculated by carrying out the

second-order approximation mentioned by Hermach in Appendix 2 of reference [4] to give, with reasonable

accuracy, 8p = -{-Ad-/Tod, where To is the heat-sink temperature. Of course. Ad can be calculated only very

roughly, but this is sufficient. For a typical 10 mA TE

142
266 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT DECEMBER

it is less than 0.01°C, and the Peltier errors, 5<p and 5p

(independent of frequency), are less than 1 ppm. This

has been verified by comparisons with Evanohm-heater

TEs which had copper lead-in wires butt-welded to the

nickel alloy wires at the glass press. However, 5p in particular could be significant in higher-range TEs.

VIII. Low- AND High-Frequency Errors

As the frequency is lowered, the temperature of a TE begins to follow the cyclic fluctuations of the square

of the heater current. If the TE is not a linear device,

the average temperature rise may no longer be the same

as at higher frequencies. Hermach's solution of the re- sulting nonlinear differential equation for the temperature rise has been verified experimentally [4], at least

for high-range TEs having large heaters in which the

heat abstracted by the thermocouple is negligible. However, for the low range TEs we are concerned with now,

these formulas may not apply. The solutions for such

cases would be even more difficult, and have not been

attempted, particularly since the thermal mass of the

bead should then also be considered. In the simpler

case, Hermach showed that the ac-dc difference is

6i = hBd^/(x}H*, where d is the diffusivity of the heater

alloy, 21 is the heater length, aj = lirf, where / is the fre- quency, and h is a factor expressing the nonlinearities.

The same relationship might be expected for the more

complicated case, but with a different numerical factor. Thus, the agreement to better than 10 ppm among the

TEs at 20 Hz, shown in Table I, with their varying

lengths, gives strong verification to the absence of significant low-frequency errors down to this frequency.

These reference TEs have heaters less than 1 by 10~' cm in diameter, with resistivities of 120 /ifl cm. The

computed errors from skin effect are much less than 1 ppm even at 50 kHz. The ac and dc conductances and

the capacitance between binding posts on the phenolic

top of the box in which the TEs are mounted are much

less than the limiting values of 0.01 nA/V and 100 pF,

respectively, which might cause 1 ppm errors at audio

frequencies.

Reactance and skin-effect errors in the resistor units

of the TVCs, computed according to formulas developed

by the authors [3], are also less than 1 ppm through 50

kHz up through the 100 volt range, and the comparisons

verify this. They cannot easily be calculated for the

higher ranges, which have several resistors and adjustable inner shields. Additional reactance errors can arise from the lumped capacitance C of the connector between the resistor and the TE. This shunts the heater,

which at these frequencies can be considered as a lumped

resistor R in series with an equivalent lumped in- ductance L. It can be shown that the resultant effect of

this capacitance on the magnitude of the heater current approaches a^/2—a0 as the series resistance (voltage

range) is increased, where a=ojCi? and (8=ajL/i?

(a and (3«1). The computed error of less than 1 ppm at 50 kHz has been verified by deliberately doubling the

capacitance.

IX. Other Errors

Peltier effects at the heater and lead-in junctions can

also cause a second-order change in the effective dc re- sistance of a TE. The calculation is difficult and the re- sults uncertain, but the change should be much less than one ppm for the 5 to 20 mA TEs in the NBS reference

group. Thus, this cannot account for the somewhat

larger changes in effective dc resistance implied by the

voltage comparisons of the TEs alone (without series

resistors) in the TVCs. Even larger and more puzzling

changes have been observed in some other TEs. There is good evidence that they may be caused by rectification at oxide films, which may persist in spite of the welded

joints, but there is as yet no direct proof of this. Dc reversal errors (A/// for the same EMF) provide a good clue in studying these secondary effects. As shown

in Table IV, AI/I calculated by Widdis' formulas

should not exceed, perhaps, 200 ppm with Evanohm or Karma, and 500 ppm with nickel-chromium heaters, even if the thermocouple is mounted off the midpoint by

5 percent of the length of the heater (which is easily

noticeable). Larger differences are, thus, a sign of other

difficulties. As Hermach showed [4], the current from the heater

to ground through the bead of an ungrounded TE can

cause heating of the bead and, thus, an error. This is difficult to compute because power dissipated in the

bead is somewhat more effective in raising the thermocouple temperature than that in the heater, and the cur- rents in the two halves of the heater are no longer identical. However, rough calculations indicate that with one

volt across the bead of TE^ in the circuit of Fig. 3, the

bead resistance of a 10 mA TE must be greater than 500

megohms if the error is to be less than 1 ppm. For the

bead materials commonly used, the resistance is a sharp

function of temperature and is much less at audio fre- quencies than with direct current. At NBS, bead re- sistances are normally measured with a 2 kHz ac bridge

with rated direct current through the heater of a TE.

The results have been verified (within a factor of 2) by

ac-dc comparisons of pairs of TEs with added resistance

between the two heaters to significantly increase the

bead voltage.

X. Conclusions

A reference group of two very different kinds of TEs

has been established. The authors believe that it is very

unlikely that the average ac-dc difference of the group

differs from zero by more than 2 ppm at audio fre- quencies. This is based on the agreement to better than

0.5 ppm of the averages of the two kinds of TEs, and on

allowance of 1 ppm for the uncertainty of the average

in the network of comparisons. The results are in excel- lent agreement with calculations, which show that the

ac-dc differences should not exceed 1 ppm. With the

precise techniques that have been developed, pairs of TEs of the group may be compared with a 3<r uncertainty of about 2 ppm without any evidence to date

(in over three years of measurements) of significant

143
1966 HERMACH AND WILLIAMS: THERMAL CONVERTERS FOR AF VOLTAGE MEASUREMENTS 267

systematic errors. This is somewhat remarkable, since

fluctuations in the best available wide-range ac and do

sources greatly exceed this, and most of these TEs have

rather large residual thermal EMFs and have temperature coefficients of EMF exceeding 1000 ppm/°C.

Adjacent ranges of the TVCs may also be compared

with the same precision to step up and down the voltage

scale. Their ac-dc differences can be determined to better than 10 ppm when used with TEs of known ac-dc

differences, again without evidence of systematic error. Alternating voltage measurements can be made to 20

ppm or better by using the TVCs or ac-dc transfer

standards with appropriate dc standards.

This accuracy and precision in the ac-dc transfer, more than ten times better than heretofore available, are good indications of the virtue of simplicity in refer- ence standards. Properly designed and constructed TEs

are rather simple devices that correspond to the desired

idealized mathematical models remarkably closely, thus

providing the agreement between theory and experiment which is so desirable in a development such as this. The agreement to better than 1 ppm between the con- ventional TEs and the new very different multijunction

NPL TEs (which should theoretically have smaller er- rors), in addition to providing semi-independent verifi- cation of the accuracy of both, indicates that even

higher accuracy may be possible in the future with these new designs. Conventional single-junction TEs could

be improved by the use of bifilar heaters and closely

spaced lead-in wires with good thermal conductance

between the two halves of the heater and between the

two lead-ins. In this way, other heater alloys could be

used.

Well-designed and well-constructed ac-dc transfer

standards can be remarkably stable over most of their frequency range. Most factors which may change the

response, such as temperature and dimensional changes,

have little or no effect on the ac-dc difference. There is already some evidence that long-time stabilities of a few

ppm can be attained in properly designed and constructed TEs and TVCs. However, we have noted an

apparently abrupt change of +5 ppm in one TE, ac- companied by a change of over 100 ppm in its dc reversal

difference. Thus, TEs should be compared frequently if the highest accuracy is desired.

Comparisons of adjacent ranges, with a step-up pro- cedure such as the one described, can be used to guard

against unexpected changes in TVCs. This has already

been verified in the design and construction of two some- what similar sets of TVCs by W. Scott, K. Ballard, and

D. Bailey of the NBS Electronic Calibration Center,

Boulder, Colo. These units incorporate large wirewound resistors of controllable reactances and low temperature coefficients, so that they do not require trouble- some forced-air cooling. They will be described else- where.

The step-up procedure is also feasible with simpler

sets of TVCs incorporating fewer plug-in resistor units

and two TEs of different current ranges. Two such sets have been built but have yet to be completely evaluated.

14

Appendix I EMF Comparator

If the thermocouple resistance Ra of TEi (Fig. 3) is much less than Ri+R^, the EMF Ei is substantially the same with alternating and direct currents applied during

the test (the procedure involves holding the voltage at the output terminals of TEi constant). Then, from the

definition of ac-dc difference, {h — h) /Id = hi. Similarly,

for TE2 {la — Id')/ Id =h, where // is the direct current

that would be required to produce the same EMF of TE2 as la- The change Da —Dd in detector Di which

occurs when / is changed from /„ to Id {Id being adjusted to keep El constant) results from the change in £2- Da — Dd may then be interpreted as a measure of the

departure of Id from the value Id' which would have

produced the same £2 as would /„. Thus, combining the

above two equations and neglecting second-order terms, we have (// — /<i)//d = 5i — 62. The sensitivity of the detector to small current dif- ferences in TE2 can be determined by shunting its heater Rk with a resistor R.'^Rh to cause a small change

in its heater current A/, noting the change in deflection

AD of Di and computing s=M/IAD {M/I = Rh/R, if the heater current of TEi is unchanged). Then, if 5 is constant over the range of deflections involved, we have

8i — 82 = s{Da — Dd), as in the text. Similarly, in the comparison of two voltage converters, a resistor i?,<Ki?m can

be added in series with TVC2 to give proportional change

in applied voltage AF/ V = R,/R„, where Rm is the TVC

resistance. Then 5=AV/VAD.

These two calibration methods must leave the current

in TEi unchanged. The change AI can often be introduced and measured more conveniently in some way

(as by changing the supply voltage) that changes the current in both TEs by a small known amount. We have

found it convenient to disconnect £1, set the voltage of P to the same value, short D2, observe the resulting AD

when / is changed, and compute 5 as before.

If only a few TEs are ordinarily used in the "stan- dard" position {TEi in Fig. 3), it is convenient to make

use of the approximate output-input relation (at a fixed

frequency) for a TE, AE/E = nAI/I, where n is dependent on / but not A7 if A/«/, and is close to 2. For each

such TE, n can be determined in advance at several

values of E and plotted vs. E, since it is quite stable. Then the sensitivity of the detector in the circuit can be

determined by inserting a small known voltage A£p«£

(with a Lindeck potentiometer, not shown in Fig. 3) in series with and opposing £2, observing the change AD

in Di and computing the factor k =AD/AEp. Then, from

this and the previous equations 81 — 62 = {Id' — Id)/ Id = {Ed' — Ed)/n2E2 = {Da — Dd)/kn2E2. £2 is easily measured with potentiometer P.

If Rn is not negligible, £1 is not constant during the

test, since the currents in Ri and R2 depend on both £1

and £2. The corresponding equations for determining

61 — 62 by means of deflections of Di then contain additional correction terms involving the resistances of the

several parts of the EMF comparator, and are too com-
268 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT

plex to be useful for routine measurements. They show,

however, that if Rn<{Ri+R2) /lOO, the simplified equations already derived are valid to a few percent of the

computed 61 — 52. They show that R12 is not critical. For

our comparator, i?i +i?2=1100 f2 and Rn<lO Q, for the

usual TEs.

For each NPL converter, Rt~ 1000 ^. It is always

placed in the TE2 position, except when the two NPL

converters are intercompared, requiring the use of the more complicated formula.

Appendix II TEs IN Parallel

By using the definitions for n and 6 of a TE (see

Appendix I) we have, closely enough for small changes

in E,

Ea — Ed la — Id nEd - 8.

In the ac-dc test, nominally equal resistors are con- nected in series with TE^ and TEt, and the two branches

are connected in parallel. The sum of the two EMFs,

Ea+Eb, is held constant. Then, with second subscripts

to identity the two TEs, we have

Eaa + Eab " Eda ~ Edb ' 0

and

(1aa — Jd(\ (lab — !db\

I 1 + nbEdb [

) \ Ida / \ Idb I

( :

I + HbEdb

^ ^ = HaEaSa + ftbEbSb

If the reactances of the two paths are equal

laa ~ Ida lab ~ Idb la ~ Id = 5,

where la and Id are the sums of the alternating and

direct currents in the two branches.

If Ea = mE and Eb = (1 — m)E,

[uam + — m)]

-la - Id = n^mba + nb{\ — m)8b.

If the reactances of the two paths are not equal, additional terms will be introduced which are difficult to calculate. However, they are zero if = «(,(! —mj,

i.e., if the TEs are matched in response. Then Op = (5a + 56)/2. Even if the reactances are unequal, the

added terms are essentially eliminated if a second test

is made with the two resistors interchanged and the

results averaged.

Acknowledgment

The authors express their appreciation and thanks to

F. J. Wilkins for the two multijunction TEs, to L. Julie

of the Julie Research Laboratories for suggesting the

basic idea of the EMF comparator, and to the following

for their cooperation in the manufacture of other TEs

used in this group: the Levy brothers of the American

Thermoelectric Company, L. R. Graham of Graham

Associates, F. Gay of the Best Products Company,

M. Rosenfield of the former Field Electrical Instrument

Company, and L. W. Pignolet of Weston Instruments

Incorporated.

The use of these particular TEs with heaters of Evanohm and Karma should not in any way be construed as constituting an exclusive endorsement by NBS of these

products or alloys. Other TEs, made to the same specifications with the same or other alloys of equally low

thermoelectric effects, should be equally suitable.

References

[1] "American standard requirements for electrical indicating instru ments." New York: American Standards Association, C.39.1

1964.

[2] F. J. Wilkins et al., "Multijunction thermal converter," Froc lEE (London), vol. 112, pp. 794-806, April 1965.

[3] F. L. Hermach and E. S. Williams, "Thermal voltage converters

for accurate voltage measurements to 30 megacycles per second,'" Trans. AIEE {Communication and Electronics), vol. 79, pt. I, pp

200-206, July 1960.

[4] F. L. Hermach, "Thermal converters as ac-dc transfer standard;

for current and voltage measurements at audio frequencies,'

/. Res. NBS, vol. 48, pp. 121-138, February 1952.

[5] W. J. Youden, "Measurement agreement comparisons," Proc

1962 Standards Laboratories Conf. Washington, D. C: NBS

Misc. Pub. 248, August 16, 1963, pp. 147-152.

[6] F. C. Widdis, "The theory of Peltier and Thomson effects in thermal ac-dc transfer devices," Proc. lEE (London), Monograph

497M, January 1962.

[7] Landolt-Bornstein, "Zahlenwerte und Funktionen aus Physik,

Chemie, Astronomic, Geophysik und Technik," 6. Auf., Band II/6. (6th ed. vol. 2, pt. 6). Berlin: Springer-Verlag, 1959.

[8] D. Flach and L. A. Marzetta, "Calibration of peak a-c to d-c comparators," Froc. 1965 ISA Conf., pt. 1, vol. 20, paper 14.

145
Section 9

JOURNAL OF RESEARCH of the National Bureau of Standards — C. Engineering and Instrumentation

Vol. 75C, Nos. 3 and 4, July-December 1971 Thermal Voltage Converters and Comparator for Very

Accurate AC Voltage Measurements

E. S. Williams

Institute for Basic Standards, National Bureau of Standards, Washington, D.C. 20234

(July 22, 1971)

A new fourteen-range set of thermal voltage converters and a thermoelement comparator are used

to measure ac-dc difference, and a-c voltages relative to external d-c standards, with 20 ppm (parts- per-million) accuracy at audio frequencies. The imprecision is less than 2 ppm. Corrections relative to the very stable middle ranges can be redetermined for every range by a seven-step intercomparison of

certain adjacent ranges. Key words: AC-DC difference; comparator; thermoelement; transfer voltmeter; voltage measurements.

1. Introduction

An a-c voltage at audio frequencies is measured

most accurately at the present time by using a thermal

voltage converter (TVC) to compare it with a stable

and accurately measured d-c voltage, which is nominally equal to it. The basic d-c standards are then,

in effect, "transferred" to the a-c measurement. The

TVC may be simply a thermoelement (TE) in series

with an appropriate multipHer resistor. The output

emf of the TE is ordinarily monitored with a nuU

detector and a balancing circuit, which may be a Lindeck potentiometer. A balancing circuit and a null detector are included in most commercial multirange models.

Before the TVC is used for a-c voltage measurements

it must be tested for ac-dc difference or frequency in- fluence so that corrections may be applied. The set of TVCs and the new TE comparator described in this paper were developed primarily for making these

tests. However they may also be used for a-c voltage

measurements as explained in section 9. For ac-dc

difference measurements the TE comparator, whose

read out instrument may be either a nanovoltmeter or a galvanometer, provides a considerable advantage over

other methods in overcoming the difficulties caused

by power supply instability and inexact voltage control.

The 14-range set of TVCs (designated No. 7) consists

of six resistor units and two TEs, and extends from 1 to 1000 V. The TEs are rated at 2.5 and 5.0 mA, and each one may be attached to any one of the resistors by a coaxial connector (see fig. 1). This permits each resistor

to be used for two voltage ranges, as shown in table

1. Certain adjacent ranges of the set can be inter- compared, and the ac-dc differences of all the ranges

can be determined relative to any one range. This

set is otherwise similar to an earlier set [1J\ which

consisted of two 5-mA TEs and 12 series resistors,

and covered a range from 0.5 to 500 V. The earlier set, designated No. 1, has been extended to 1000 V with one additional resistor as part of the present project.

It is evaluated in a 14-step intercomparison in which

each range is compared, at reduced voltage, with the

next lower one.

In a well designed TVC the ac-dc difference is not affected by changes in the applied voltage. Therefore, an ac-dc difference determined for the 100-V range at 60 V can be applied as a correction when this resistor

is used at 200 V to test the nominal 300-V range of the next higher resistor. Experience has shown this stepup procedure to be feasible. However if the higher volt- age ranges are not carefully built the ac-dc differences can change due to self-heating effects to be described

later.

2. Design and Construction

The two TEs are mounted in 2-in brass tubes

(see figs. 1 and 2, upper right) with coaxial connectors

for attaching to a resistor. The 2.5 mA-TE has a 400 Cl heater and is used alone as the 1-V range. The 5 mA-TE

has a 125 fl heater and a 275 fl resistor is added inside

its enclosure to make a 400 fi, 2-V range.

The middle ranges (six ranges, 3 to 60 V) make use

of three resistors — 0.8 kCl, 3.6 kii, and 12 kfl (see

table 1). Each of these is a 2W metal-film resistor

mounted coaxially in a 3-in brass cylinder 4 1/2 in long with coaxial connectors at each end.

' Figures in brackets indicate the literature references on page 153.

146
Figure 1. New set of thermal voltage converters. The simple symmetrical geometry of these units

permits approximate calculations of their reactances

[2]. The calculations are inexact because of the nec- essary assumptions as to end-effects, and they neglect

any small residual reactances of the resistors themselves. They do indicate however, that the frequency

error of each resistor unit (without the TE) should be

less than 1 ppm at 50 kHz, even for tubes smaller than

2 in in diameter. (The 3-in tube was chosen for the

higher ranges where more space is necessary and was

therefore used for the middle ranges also.)

The higher voltage ranges have larger resistor as- sembhes to avoid excessive temperature rise. The

100-200 V unit has a 40 kfl spiraled metal-film resistor

five inches long.^ The 300-600 and 500-1000 V units

have resistors of tin oxide deposited on a 0.81-in glass

tube 6 in long. A small voltage coefficient was antici- pated in these resistors, but measurements have shown

it to be negligible.

^This experimental resistor is not ordinarily available commercially. Four 40 kfl metal

film resistors in series-parallel can also be satisfactory.

Table 1. NBS thermal voltage converter set No. 7

Fourteen voltage ranges (column 2) are formed with six series

resistors and two TEs (F 1 and F 2).

Errors in these ranges are caused mainly by capacitance between the resistor assembly and the outer

casing, which permits alternating current to bypass

the TE to ground. Therefore more a-c than d-c voltage

is required for a given TE output. Frequency compensation could be provided by placing relatively small

capacitors in parallel with part of the resistance.

However, such capacitors probably would not be suf- ficiently stable over a long period of time and might

be affected by temperature changes which occur in the

resistor enclosure.

Compensation was therefore provided (as in the

earlier set No. 1) with an inner shield which is con- nected to the input and surrounds the high- or inputend of the resistor (see "S" in fig. 2). The shield is positioned, relative to the resistor, to control the capacitance currents and provide optimum high-frequency

Series Voltage TE Total

resistor range resistance AH mA AO

1 2.5 (Fl) 0.4

2 5.0(F2) 0.4 3 2.5 1.2 0.8 6 5.0 1.2 10 2.5 4.0

3.6 20 5.0 4.0 30 2.5 12.4 12 60 5.0 12.4 100 2.5 40.4 40 200 5.0 40.4 300 2.5 120.4 120 600 5.0 120.4 500 2.5 200.4 200 1000 5.0 200.4 Figure 2. Essentials of thermal voltage converters.

ThermoelemenI (TE) enclosure, low voltage unit with resistor (Rl. and hi^di vnliajie unit ( 100 to 1000 V) with resistor (R) and frequency compensation shield (Si.

147
compensation; perfect compensation is possible only

at one frequency.

3. 1000-VTVC

As stated earlier the corrections for the high and low

voltage ranges are determined relative to the middle

range by intercomparison tests. It is necessary therefore that the ac-dc difference of all ranges be unaffected by voltage level. The voltage effect is rare in low

and middle ranges, but it can be troublesome at higher

ranges where the heat generated by the resistor is appreciable. All intercomparison tests were initially

made at two voltage levels in order to evaluate this effect — or to prove its absence. If, for example, the

nominal 600-V range is compared with the 1000-V

range at both 400 and 600 V, and the same results are obtained, it is unlikely that the 600-V range is affected

by the voltage increase (the 1000-V range is very

probably unaffected at these voltages). However both

ranges are in question until we are assured that the

1000-V range is also free of this effect. The highest range, then presents a special problem.

It is more likely to change due to increased voltage

(heating) and there is no higher range to compare it with. If it is compared with another 1(X)0-V TVC at two or more voltages and no change is observed it is possible that both have the same voltage effect. However this is unlikely if TVCs are of somewhat

different design. If intercomparisons of several units

of difference design show no change between rated

voltage and a lower voltage we may safely assume

this effect is negligible in each one. Five 1000-V TVCs (table 2) have therefore been

built, each with differing components, and a network

of intercomparison tests was made as described in the next section. The original set of TVCs [1] was included

in these tests. Table 2. Five 1000-V WCs

Unit Resistance yoltage range with TE Resistor assembly

F 1 F2

A

kn

400 1000 Four loom wire- wound resistors In series, each 1 3/8 in long.

B 400 1000 Four loom wire- wound resistors in series, each 1 3/8 in long. C 200 500 1000 Four 200 AO metalfilm resistors in series-parallel, each

2 in long. D 200 500 1000 One 200 mtin oxide

resistor 6 in long.

E 400 1000 Four 400 Afi metalfilm resistors in series-parallel, each

2 in long.

The five 1000-V TVCs are designated alphabetically

(A through E)^ and all except "A" have the inner shield, as described below, for frequency compensation. The

resistor in unit "A" is mounted concentrically in a set of six brass rings whose potentials are maintained by a capacitance divider. The whole assembly is mounted in a 4-in brass cylinder 7 in long.

Self-heating of a TVC may affect either the fre- quency compensation or the resistor itself. Apparently

the dielectric losses in the insulation between con- ductors can be affected by heating and change the

impedance of the resistor. This change occurred within

a few minutes after a voltage increase or decrease in several resistor types, which "were therefore discarded.

The frequency compensating shield may be moved,

relative to the resistor, by thermal expansion of mounting parts and cause a change in ac-dc difference. This

occurred in some of our early units and the change was slow, taking up to an hour. Later shields were

therefore mounted very rigidly since a small displacement will have a large effect on the frequency influence.

The shield now in use is in two parts. A brass

cylinder, with one end closed, is mounted firmly

against the outer end piece (input end) with a polystyrene insulator one-half inch thick (fig. 3). A movable

tube fits tightly inside this cylinder and the shield

length adjustment is made by pushing this piece for- ward with a small rod inserted through the end piece

and the insulator. The rod is removed after each

adjustment.

The shield is most effective at the leading edge

where the voltage difference is greatest between the

resistor and the shield. Therefore the movable part of

the shield is cut at an angle, so that only part of it (see fig. 3) extends outside the fixed cylinder. Thereby

the effect of small position changes is reduced. Even

so, the compensation is affected by approximately 0.01

percent per millimeter of position change in some

typical units.

The shield length adjustment is made in small steps,

and tests are made after each adjustment. When

optimum compensation is achieved, the unit is opened

and the shield parts are locked together with three screws through their overlapping portions.

4. TE and TVC Tests

As stated earlier the middle ranges are the most

accurate, and they are the base from which higher

and lower ranges are tested. However they are no more accurate than the TE used with them. Before

being installed in a TVC the TEs are therefore compared with one of a group of carefully selected TE standards which were made according to NBS specifications.

This group contains twelve elements made by four

manufacturers, and are rated from 5 to 20 mA. [1] They

have either Karma or Evanohm ^ heaters to reduce

' Unit "D" is assigned to this set (No. 7).

* Certain commercial products and instruments are identified in this paper in order to specify adequately the experimental procedure. In no case does such identification imply recommendation or endorsement by the National Bureau of Standards, nor does it imply

that the products or equipment identified are necessarily the best available for the purpose.

148
Figure 3. Cross section ofhigh voltage TVC.

Inpui end is at the left. Fixed and adjustable shield parts are supported by a polystyrene insulator one-half inch

thick.

thermoelectric eflfects in the heaters. They also have

small reverse d-c differences (less than 200 ppm),

hio;h bead resistances (over 1000 Mfl), and small ac-dc

differences. An additional two are multi-junction

elements of different design. All of these TEs have

been intercompared as current converters with agreement to better than 3 ppm. Theoretical calculations,

taking into consideration all known sources of error, also indicate that the error should not exceed a few

ppm at audio frequencies. Because of this and because

of the excellent agreement between these thermoelements of different design and construction, we

may safely assume that the average ac-dc difference

of the group is 0 to ± 2 ppm.

The low-frequency performance of a TVC depends

on the TE since the reactance of the resistor is entirely

negHgible even at a few kilohertz. The accuracy of

a TE at low frequency is mainly dependent on the

length of the heater. A very short heater permits heat

to flow more readily to the support stems and coohng

to occur between current peaks. However most commercial TEs have errors less than 10 ppm at 20 Hz

and some have errors less than 10 ppm even at 5 Hz.

In testing selected commercial TEs for the TVCs

at audio frequencies we often find a small ac-dc

difference (up to 4 ppm) which is independent of

frequency but dependent on heater current. It or- dinarily decreases at lower heater currents. When

these TEs are used in a TVC the ac-dc difference

will therefore be slightly voltage dependent. In the

step-up test process, where the TVC is alternately

used at 50 and 100 percent of rated current, this rela- tively small thermoelement error is introduced at every step of the process. If the process is a long one,

as in the 12-resistor set, there could be a significant

accumulation of error. TVCs in the new set are inter- compared by testing each resistor once with one TE

at reduced current (60 or 67%), and again with rated

current on the other TE. If these TEs differ in ac-dc

difference at these currents, there will again be an accumulation of error. This is one of the advantages of the set with only

six resistors. If the 20-V range is the starting point

(it is probably the most stable), we have only four

steps up to the 1000-V range, and three steps down

to the 1-V range. The other advantages are that less time is required for the intercomparisons and there

are fewer units to construct.

Intercomparison data for this set of TVCs are listed

in table 3 as well as determinations made using TVC

set No. 1. (Test methods are discussed in later sections

of the paper). The middle ranges differ by 2 ppm or less even though the resistance varies from 1200 to 12000 H.

Since the resistance enters into the theoretical calculation of ac-dc difference, this agreement is an additional indication of accuracy.

Step-down tests show that the two lowest voltage

ranges usually have a positive correction (i.e., more

a-c than d-c voltage is required for a given TE output)

of 5 ppm or less at 50 kHz. Ther^ is no resistor in series with the TE at the 1 V range, and at the 2-V

range the TE is a large part of the total impedance.

These ranges are therefore affected more by the small

reactance of the TE. The correqtions for these ranges

are more likely to change if a TE is replaced than for the higher ranges. A step-down test from the middle

range is therefore advisable when TE replacements

are made.

The five 1000-V TVCs were compared with the

300- and 500-V ranges of TVC set No. 1 and the 600-V

range of the new set No. 7. The corrections are listed

in table 4 for 20 and 50 kHz and all values are averaged.

Table 3. AC-DC difference

TVC-a TVC-b

Applied

voltajie

AC-DC difference (ppm)

20 kHz 50 kHz

8« 5 5,-6/, fix 6

Fl 1 -3 -3 + 1 -3

F2 Fl 1 4-

1

0 F2 2 -2 -2 + 1 -2

Fl-3 F2 2 + 1 - 1 Fl-3 3 -1 + 1 0 -2

F2-6 6 -1 0 0 -3

Fl-10 F2-6 6 + 1 + 1 Fl-10 10 0 + 1 + 1 + 1 F2-20 20 0 0 + 1 + 1 Fl-30 F'2-20 20 - 1 -2

Fl-30 30 - 1 +

1

-1 + 2 F2-60 60 - 1 0 -1 0 Fl-lOO F2-60 60 -1 -4

Fl-lOO 100 -2 0 -5 -2

F2-200' 200 -2 -

1

-5 -6

F 1-300 F2-200 200 -2 + 6 F 1-300 3(X) -4 -3 + 1 +4

F2-600 600 -4 -4 + 1 + 1 F 1-500 F2-600 500 -2 + 26 Fl-500 500 -6 -3 +27 +30

F2-1000 1000 -6 +27

149
Table 5 lists intercomparison data for the same fre- quencies for seven pairings of the five 1000-V TVCs.

Values for 600 and 1000 V do not differ significantly and

so they are all averaged together.

Corrections (8) for each range of the new TVC set are listed as determined in comparison with TVC Set

No. 1. Intercomparison data(8„ — S^) are also listed and

a computed correction (8.s) based on the intercomparison, starting with the average of the measured ac-dc

differences of the 10- and 20-V ranges. The TVCs are

identified as the combination of a TE (Fl or F2) with

the appropriate resistor; i.e., F2-600 forms the 600-V

range.

Averages of the corrections for the five 1000-V

TVCs listed in table 4 are shown in the circles in figure

4, and the connecting lines show the intercomparison

values from table 5. The agreement between the vari- ous units is illustrated by this figure.

Each of the values listed in tables 4 and 5 is the

average of at least four separate determinations.

An analysis of the data taken in a series of 26 inter- comparison tests (an average of 5 determinations

with each one), involving all ranges at two frequencies and usually two voltages, showed an imprecision

(3 o-Q, where o"a is the standard deviation of the aver- age of five determinations) of only 1.4 ppm.

Table 4. Ac-dc difference corrections for five lOOO-volt TVC (A-E)

The standard instruments are the 300- and 500-V units of Set No. 1 and the 600-V range of

Set No. 7.

Standard Volts A B C D E

AC-DC difference at 20 kHz (ppm)

.SOO-1 500 -20 -9 -4 -12

.•^00-1 300 -4 -10

600-7 600 -3 -22 -2 -12,-10

-8 -3 -10

Ave. -6 -21 -9 -3 -11

AC-DC difference at 50 kHz (ppm)

500-1 500 + 21 -6 + 11 + 30 -3

300-1 300 + 30 -4

600-7 600 + 30 -9 + 30 -6

+ 23 + 32 -6

Ave. + 25 -7 + 11 + 30 -5

Table 5. Relative ac-dc differences for seven pairings of members of

the five 1000-V TVCs

Volts Sa-8b 8b- 8c 8c-8o 8|)-6e 8a-Sc 8a-8d

AC-DC difference at 20 kHz (ppm)

600 -11 + 14 + 2 -7 + 7 + 3 0

1000 -17 + 13 0 -6 0 + 9 + 5 600 -16 + 17 - 10 -2

1000 -17 + 5 Ave. -15 + 15 + 1 -7 + 4 + 6 + 2 AC-DC difference a t 50 kHz (ppm)

600 -20 + 14 + 10 -33 + 28 -15 + 3

1000 -22 + 6 + 2 -33 + 35 -14 + 4

600 -25 -37 -15 + 2 1000 -24 + 1 Ave. -23 + 10 + 6 -34 + 32 -15 + 2

5. Thermal Compensation

The coaxial connect()r between the TE and the series

resistor provides a low-reactance connection for minimum frequency error. Tests and calculations show the

error is less than 1 ppm at 50 kHz. However, it also

permits heat to flow freely from the resistor, which

dissipates up to five watts on the 1000-V range, to the

TE which has a temperature coefficient of emf of ap- proximately 0.2 percent/" C (at constant input current).

Heat conduction through the center conductor of

this connector was reduced, at 300 V and higher, by

replacing the two polystyrene insulators in the coaxial connectors (one in the resistor output and one

in the TE input) with two made of boron nitride. This

material is a good electrical insulator and yet it con- ducts heat well enough to provide an effective heat

sink for this conductor. It is used in both ends of the

high voltage resistors to improve the heat flow from the

resistor to the relatively heavy brass casing. This also

helps to reduce the temperature rise in the resistor. Heat flow through the outer part of these connectors

will gradually raise the temperature of the TE casing,

and this also causes a heating drift. The increase in temperature of the TE bulb is slowed by mounting it in a short section of brass tubing to increase the

thermal mass and this tube is thermally insulated from

the TE casing. Thermal compensation is also added by

attaching a lO-O thermistor to the tube containing

the TE and connecting it in series with the TE output.

A resistor appropriately chosen to complete the compensation circuit (see Fig. 7 and appendix I) will draw current through the thermistor sufficient to compensate for the temperature increase.

This resistor value may be computed for TEs with

average temperature coefficients as in appendix I. However, temperature coefficients differ considerably

and, for best results, it is usually necessary to match

the resistor to the TE and thermistor combination. A

suitable value can be found by substituting a variable

resistor outside the TE casing and adjusting it for minimum warmup drift on a 1000-V range. However, if the

TVC is to be used with a TE comparator similar to the one to be described, it will frequently be connected

20 l<Hz 50 kHz

Figure 4. Intercomitarisons of lOOO-volt TVCs.

Ac-dc difference c<trrecti<»ns (ppm) from table 4 for five lOOO-V TVCs are shown in the circles and the relative differences from table -S are shown on the interconnecting

lines. The algebraic difference between any pair should be equal t<i the figure on the connecting line within experimental error.

150
across a lOOO-O divider, and this will affect the compensation by about 20 percent. A value about 10 per- cent higher than the one chosen experimentally should

provide a satisfactory compromise and reduce the

drift by a factor of five or more. With the compensation

and the large thermal time constant of the TE in its casing, the drift can be reduced to less than 5 ppm/

minute under typical laboratory condition. The error from this is almost completely eliminated by taking a sequence of readings with a-c, d-c, reversed d-c, and a-c voltage applied, as described in the next section.

6. TVC Test Methods

A brief review of other comparison methods may be

useful before describing the TE comparator. Figure 5

shows two TVCs connected in parallel. The separate

balancing circuits (or Lindeck Potentiometers) (B) and

null detectors (D) may or may not be built in. Stable a-c and d-c voltage supplies are connected to the two TVCs

by the ac-dc switch. The switching arrangement shown

is typical and convenient, although others, such as re- versing the d-c supply by a separate switch, are satis- factory. The d-c supply must have an ungrounded

output and it should be possible to ground either ter- minal as the voltage is reversed. The reversed d-c dif- ferences of the TVC can rarely, if ever, be neglected

in these tests. For convenience and clarity, we shall call one TVC

the "standard" and the other the "test." Normally the one with known ac-dc difference corrections is the

standard, and the test TVC is usually treated differ- ently in the test procedures, as explained below.

There are two procedures for this intercomparison.

One might be called a null-balancing method. With a-c voltage applied to both TVCs, the balancing circuits of

the test and standard instruments are adjusted to null the detectors. The d-c supply is then switched to the

TVCs, and the voltage is adjusted to obtain a null bal- ance first on the detector of the test instrument and

then on the standard. Each d-c voltage is carefully

measured, and the small difference between the two

should equal the difference between the a-c voltage and

one direction of d-c voltage. The measurement is then

repealed with reversed d-c voltage, and an average is taken.

This method is tedious and time consuming, an ac- curate d-c voltage standard is required, and if there is any drift in either instrument output due to warmup or other reasons, as there very often is, the data may lack

both accuracy and precision.

A more satisfactory procedure and one that will minimize the effect of drift may be called a deflection

method. Both balancing circuits are adjusted to near

null, and then a-c and d-c voltages are applied in the

succession indicated in figure 5 (AC, DC+, DC—, AC).

For each input the power supply is adjusted to set the

test instrument detector to null. The deflection of the

standard TVC detector is read for each setting, and the

difference between the average reading on a-c voltage

and d-c voltage is an indication of relative ac-dc differ- ence. If the readings are made at nearly equal time

intervals, and if the drift rate is reasonably constant

(even though fairly large), the determinations can be

made with satisfactory precision.

If the detector of the standard TVC is a galvanometer

the reading may be in divisions or millimeters, but they can be converted to volts or percent ac-dc differences

by calibrating the detector scale in one of several ways.

A calibration factor can be determined simply and di- rectly by making a small measured change in the d-c

input and observing the detector response.

If the detector is a microvoltmeter and if the balancing circuit is a Lindeck potentiometer indicating the

millivolt output of the thermoelement, the calibration

step can be eliminated. With the four deflection readings on the detector the difference between the average

reading on a-c voltage {£„) and those on d-c voltage

(Eii) is computed from the differences of the corresponding readings of the microvoltmeter. The ac-dc differ- ence of the test instrument is

^ . Esa Esd

where the subscript s indicates that these values come

from the standard instrument, and 8s is the ac-dc differ- ence correction for the standard.^

The factor n relates small changes in thermoelement

heater current (A/) (at a fixed frequency) to corresponding changes in output emf (A£'/£'= nA/// approximately). For a TVC we may substitute n^VIV for the right

hand expression, so that n= VAE/E^V. The value of

/I is 2 if the thermoelement has a square law response,

possibly at very low heater currents, but it is usually

1.7 to 1.9 at rated current. Determinations of n should

therefore be made in a d-c test at five or more current

levels by measuring the values indicated. A plot of n against E can be made so that values corresponding to any emf can be found for substitution in the equation

above (see sec. 8).

Fl(;URE 5. Two-Potentiometer method for TVC comparisons. ^ See eq 3 of appendix II. This general formula is applicable here also.

151
7. TE Comparator

There are several variations of basic comparator

circuits which are used to minimize the difficulty

caused by power supply instability [1, 3]. They employ

a voltage divider circuit to which two emfs are con- nected. The divider is adjusted to null a detector,

and at this point the divider setting corresponds to the ratio of the emfs. If one emf {Et) is held constant

as a-c and then d-c voltage is applied to the TVCs,

the other emf {Es) will change if there is a relative

ac-dc difference in the TVCs at the frequency {8i ¥^ 8,). The resulting unbalance in the divider will produce a change in the detector deflection proportional to the

ac-dc difference. However, small fluctuations in the

power supply will produce nearly equal proportional

changes in the emfs and therefore the detector will not be affected appreciably. The stabilizing effect depends on how well matched the time constants

and response characteristics of the TEs are, but the

effect usually affords a significant advantage. Also,

the monitored emf need not be held constant so

exactly as with a simple balancing circuit. In the new comparator (fig. 6) the higher of the

two emfs is connected across a lOOO-d Kelvin-Varley

divider, and the lower one is connected to the variable

tap. The detector is brought to null by adjusting the

divider. A Lindeck potentiometer and the same de- tector (with key k\) is used to monitor the test TVC

output {Et). The detector is labeled "N" in the dia- gram because the use of a nanovoltmeter is suggested.

The test procedure is similar to that already mentioned. Preliminary settings of the potentiometer

and divider are first made so the readings of the

nanovoltmeter will fall near mid-scale. Then with a-c voltage applied, the key ki (labeled "SET" on the

instrument, figs. 8 and 9) is depressed and the appro-

+oc oc

priate power supply is adjusted for a null on the

detector. Key k2 (READ) is then depressed and the

detector indication is read. The two-step procedure

is repeated for two directions of d-c voltage and for a second a-c input. The average indication in nanovolts ** (M,) for a-c input and the average for d-c input

(Nd) is computed and the ac-dc difference of the

test instrument is

8, = 8,s +

TlsEs (appendix II eq 5)

Figure 6 shows the test TVC output (£",) connected

to the high emf terminal (E// in fig. 8). If Et is smaller

than Ex it will be necessary to interchange them, but

this is a minor inconvenience. If Et is connected to the E| terminal the potentiometer is switched to that

input with switch S2 (fig. 8) and Et is monitored as

before. The same switch also reverses the polarity

of the detector, and therefore the sign of the indication. The slightly modified equation for 8; is eq 11 in appendix II. A galvanometer may also be used as a detector but

since the circuit resistance is relatively high a photoelectric amplifier is necessary for sufficient sensitivity.

The scale must be linear over the portion to be used

and a scale calibration is also necessary to relate the

readings in divisions or millimeters to a voltage change.

The calibration can be made quite easily however by

moving switch Si from READ to CAL with k2 closed

and observing the resulting deflection change. This

increases the resistance of the 1000-fl divider by 0.5 (1 and changes the divider current by 0.05 percent. The

^, READ : :0.5a

: 0

• pot"r5 ~~~^t—

;

I.I kO.'.

c— —

;200n

1

Eh

I— nTEST

SET READ

Figure 8. New TE comparator.

Figure 6. New TE comparator. 6/ — 6a is conveniently in parts-per-ntilliun if E, is in millivolts and Nh — N„ is in nanovolts.

152
Figure 9. Photograph of TE comparator.

voltage applied to the galvanometer circuit is changed

by the same amount and the deflection change Di

is used in eq 10 and 12 in appendix 2 to determine

di. A measurement of Es is made only to find the cor- responding value of Us from a plot as before.

8. n Tests

Measurements of n are conveniently made v^^ith the

comparator. The thermoelement to be tested is con- nected to the El input and the Eh input is unused. A

key labeled "n TEST" (fig. 6) is locked dovk^n and

switch S2 is moved to Eh connecting the potentiometer

voltage across the divider. The divider is set arbitrarily but preferably near full range and the potentiometer is adjusted to null the detector. The input

voltage is then changed by a small measured amount

M'lV, and the resulting change in TE output (^E)

is measured by observing the change in indication of

the detector (nanovolts or microvolts).^ Switch S2

is then moved to Ej and a measurement of TE output

(£") is made with the potentiometer. The values for each factor of the equation given above [n=Vb£j

EAl ) are then known and n is calculated from this equation.

If a galvanometer is used as the detector a measure- ment of Di is made as described above, and the deflection change resulting from the measured change in input voltage (AF) is designated D>. Then n = 0.05

DJqDi, where q is the percentage change in input

voltage (100 AVIV).

Determinations of n need be accurate only to a few

percent, and, since they are quite stable, retesting is not usually necessary.

9. AC Voltage Measurements

As suggested earlier, the comparator may be used

with one TVC to measure an a-c voltage such as the

A small I hangp in ihr iiipul !> a TVC can be measured with a voll b.n and piilenli

i)nit-l<-r or by a si-llinj; witli a calibrated d c milage supply. It is also i nnveiiieni to use a

resistiir with a shurlint; switch in series with a T\ CI ti> inlmduce a small current < hant;e {U) in the thermoelement heater (i.e., 20 SI with a 40 k!l. 200 V TVC will give 100 A///

0.05 percent which is eijuivalent In lOOAVlV).

input to a digital voltmeter or the output of a calibrated

a-c voltage supply. Switch Si has a third position

marked "POT'R" which disconnects the divider from

the Eh input. If a TVC output is connected to Eh, and

switch S2 is moved to that input, the potentiometer and

detector can be used to measure the TVC output and

changes in it. The circuit is as figure 5 without the test TVC. The d-c voltage is accurately measured and nominally equal to the a-c voltage.

It is suggested that d-c voltage be applied to the TVC

first and a detector reading taken. Then, at nearly equal

time intervals, a second reading is taken with a-c volt- age and a third with reversed d-c voltage. The differ- ence between the average of the readings with a d-c

voltage [Ed) and the one with a-c voltage (£"0) is related

to the input voltage difference very much as before.

With a nanovoltmeter the emfs are read directly, but

if the detector is a galvanometer a scale sensitivity

factor must be determined.

10. Design Details

The case of the TE comparator has thermal insulation and electrostatic shielding, and the comparator is carefully built to minimize thermal emfs and contact

resistances. The switch decks of the Kelviii-Varley

divider and Si and S2 have enclosed silver contacts and

the keys ki and k2 are of a low-thermal type. The back

contacts on these keys connect a lOO-H shunt across

the detector to reduce sensitivity and noise until one

key or the other is depressed. The fine control tor the

divider is a ten-turn helical resistor with a special lowthermal sliding contact. The resolution of the uncalibrated divider is not quite sutticient for setting an exact

null on the detector, but setting within a few divisions

is adequate in the ordinary use of the instrument.

153
11. Conclusion

The use of one resistor unit for two TVC voltage

ranges has reduced by about half the time required, over previous TVC sets, to make a complete intercomparison test. Changes in any member of the set are

readily detected since frequent intercomparison tests are more feasible.

The TE comparator combines the stabilizing fea- ture of other comparators with much of the simplicity

of the single Lindeck potentiometer or balancing

circuit. It is relatively inexpensive to build and calculations are simpUfied and direct, especially where the

detector is a nanovoltmeter. Repeated tests have shown

that the imprecision is less than 2 ppm and ac-dc transfer accuracies, with the TVC set, are 10 ppm at audio

frequencies and 20 ppm up to 50 kHz.

change in thermistor resistance, ARt—aRtI 100

where a is the temperature coefficient of the thermistor

in percent/" C — typically about —4 percent/" C. Therefore, for thermal compensation,

p=100iiR,IR = aR,IR

and R = aR,I (3

With the typical values hsted we find R = 200 ft, and, subtracting Rc and Rt, the value for Rg is 180 ft.

14. Appendix II

Development of Equations"

The ac-dc difference of a TVC is defined as

This work was supported in part by the Army Metrology and Calibration Center, Redstone, Alabama.

Their encouragement and support are gratefully

acknowledged. The author also acknowledges the

help of C. B. Childers and A. G. Perrey who made

many of the exacting measurements described here.

12. References

[1] Hermach, F. L. and Williams, E. S., Thermal converters for audiofrequency voltage measurements of high accuracy, IEEE

Transactions on Instrumentation and Measurement, IlVI-15, No. 4, 260-268, (Dec. 1966).

[2] Hermach, F. L. and Williams, E. S., Thermal voltage converters

for accurate voltage measurements to 30 megacycles per second.

Trans. AIEE (Communication and Electronics), 79, Pt. I, 200-206, (July 1960).

[3] Turgel, Raymond S., A comparator for thermal ac-dc transfer standards, ISA Transactions, 6, No. 4, 286-292, (1967).

13. Appendix I

The temperature coefficient of emf, /3, of the

average TE is about —0.2 percent/" C, with constant

input current. Thus a corresponding decrease in the

resistance, R = Rc + Rt + Rs (see fig. 7) is required

to maintain a constant voltage across the resistor

Rg. For a 1 degree increase in temperature, the

Figure 7. Thermal compensation for TE.

Kr TE oulpui resistance (about lOfl). H, Thermistor resistance (about lOfl), Rt Series resistor to complete compensation circuit. {Rr and R, have negligible temperature coefficients.)

8 =

Vd

(1)

where Va is the a-c voltage and Va the average of the

two directions of dc-voltage required to produce the same output emf. In the circuit of figure 6 the a-c and d-c voltages are adjusted to give the same emf

Et, of the TVC under test, as indicated by a null on

the detector with ki closed. The same voltages are appHed to the standard TVC. Then,

Va=Va{\ + bt) = V'a{\ + ^s) (2)

where V'^ is the d-c voltage required to produce the same output emf of the standard as Va, and the subscripts, t and 5, refer to the test and standard instruments respectively. If Vq— Vg « 1, then, closely

enough

8f — 8s

From the definition of n given in the paper

Esa Esd

8i — 8s

risE;sd

With the polarities as shown in figure 6,

N=mEt-Es

(3)

(4)

where N is the detector voltage with k2 closed and

ki open, and m is the divider ratio (with the detector

resistance Ra >> Ri).

Thus.

8, = 8,+ Nd-Ng

UsE.sd (5)

where the subscript a and d have the same meaning

as before.

' These equations were developed by F. L. Hermach.

154
If a galvanometer is used as the detector, the current through it with k2 closed is, by Thevenin's Theorem,

^96 —

mEt — Es

Re

(6)

where Re is the resistance of the galvanometer circuit

with Et and Es = 0. With Ei and held constant and

pRi inserted by means of Si,

mEtl{l+p)-Es

r:

(7)

where R'^ is the resistance of the galvanometer circuit. Since mEt — Es«Es and p << 1, we have to a

sufficient degree of approximation.

/ EsP_{Dt-Dc) _Dr

tgc Re

(8)

where Dt, and Dc are the resulting galvanometer deflections and S is the galvanometer current sensitivity. In the ac-dc test, with mEt constant.

Iga lad

'sd

Da- Da

gdRe S

Thus, from (3), (8) and (9),

p (Da-Da)

igDi

(9)

(10)

The test and standard TVCs may be interchanged

if E( < Eg. If Et is applied to the El input of figure 8 and

held constant, with the detector and potentiometer con- nected as shown in the figure, a similar analysis leads

to the following equations:

8, = 8s +

TltEt

and

p{Da-Da)

n,Di

The characteristic n is determined from

^E|E

AVIV

(11)

(12)

(13)

by applying known changes in input voltage and

observing the changes in output emf with a high resist- ance voltmeter, as described in the text. If a galvanometer is used as the detector instead of the voltmeter

its sensitivity is determined by inserting pRi as described. From equations similar to (8) and (9)

AEs pAD

Es ~ D,

(14)

If the thermocouple resistance of the test TVC,

Rtc, is significant, p in equation (7) should be replaced

by p', the fraction of Ri + Rtc inserted by Si* Similarly

if the potentiometer resistance Rp is significant, p

in equation (14) should be replaced by p", the fraction

of Ri + Rp inserted by Si.

(Paper 75C3&4-321)

* with a high impedance detector. With

a galvanometer the formula for 6^ is

too complicated to be useful.

155
Section 10

Thermal Voltage Converters for Accurate

Voltage Measurements to 30

Megacycles Per Second

F. L. HERMACH

MEMBER AIEE

Synopsis: Thermal voltage converters, each consisting of a resistor in series with a

thermoelement in a coaxial line, have been

developed for measurements of rms voltages

of 1 to 200 volts at frequencies from 3 cps

(cycles per second) to 30 mc (megacycles).

An accuracy of 0.1% or better may be obtained by a-c-d-c transfer techniques up to at least 10 mc and 0.2% at 30 mc. ATHERMAL VOLTAGE converter, according to ASA Standards,' is a thermoelement of low cturent input rating

with an associated series impedance or transformer, such that the emf (electro- motive force) developed at the output

terminals gives a measure of the voltage

applied to the input terminals. Thermal

voltage converters containing thermoelements in series with wire-wound reE. S. WILLIAMS

NONMEMBER AIEE

sistors can be used to make highly acctu-ate voltage meastu-ements at audio

and ixltrasonic frequencies.*'' The fre- quency range of such voltage converters

is limited primarily by the residual re- actances of their wire-wound resistors, but tests have indicated that for some of them good performance might be ex- pected to frequencies approaching 1 mc.

This paper describes thermal voltage converters of low and computable re- actance that have been developed at the National Biu-eau of Standards to meet the need for determining the frequency limit of such instnunents.

Single-range converters with depositedcarbon resistors have been construc- ted with ranges of 1 to 200

volts. Each has a frequency influence

less than 0.1% to 10 mc and less than

0.4% at 30 mc. These rms voltage converters may also be used to calibrate 1/2%

r-f (radio-frequency) thermocouple volt- meters which are now commerdally available, and, with sine wave generators, could be used to calibrate electronic volt- meters as well.

Description and Construction of Converters

The r-f thermal voltage converters make use of the transfer principle,* in which a direct voltage is substituted for the alternating voltage to be measiu-ed. The direct voltage is adjusted to give the same output emf of the thermoelement

that was obtained with the alternating

Paper 60-135, recommended by the AIEE Indicating and IntesratioK Instruments Committee and

approved by the AIEE Technical Operations Department for presentation at the AIEE Winter

General Meeting, New York, N. Y., January 31-

February 5, 1960. Manuscript snbmltted October

30, 1959: made available for printing December 9, 1959.

F. L. Hbruacb and E. S. Williams are with the

National Bureau of Standards, Washington, D. C. The authors wish to acknowledge the help of R. Richardson and J. Scofield, who checked the

equations and carried out the compwtations which made possible Pig. 4 and the last column of Table

II, and of J. B. Hill, who assisted with some of the measurements. Thanks are also extended to M.

C. Selby and L. Behrent for their careful calibra- tion of the 1-volt converter.

156
TE

Fig. 1 . Thermal voltage converter

R = Film-type resistor 1/8 in. diameter, 2 in. long TE =Thermoelement A = Coaxial input connector, GR874PB

B=Two-pin output connector, ANIOSL

Di, Dj = Circular brass disks, 3/16 in. thick, secured to F with 4 2-56 brass screws

F = Cylindrical brass shield, 2 in. diameter, 4.25 in. long

G=Cross wire at center of Di (see text) Note: Plane of thermocouple leads at right angles to axis of F

voltage applied, and is meastired with a

potentiometer or other suitable means. Thus only good short-time stability, high

precision of reading, and small known fre- quency influence are required of the

transfer standard. Each converter consists of a cylindrical

deposited-carbon resistor in series with a

uhf (ultrahigh frequency) thermoelement

having a short straight heater in line with

its supports, coaxially mounted in a brass

cylinder, as shown schematically in Fig.

1. The residual reactances are much

smaller than those of wire-woimd resistors, and can be computed at least approximately, so that the frequency errors can be estimated. The frequency errors

of the uhf thermoelements as current converters are believed to be small to over

100 mc. The dimensions of the converters and

additional data are given in the caption of

Fig. 1. Each converter is assembled by

TE

Fig. 8. Voltage converter with compensating

shields

Si = Cylindrical brass shield, 2 in. diameter with one brass end-plate

Si=CylindricaI brass shield, 2 in. diameter

both ends open, axially adjustable (see text) Other symbols same as Fig.1 except that F is 2.5 in. diameter and 7.5 in. long Note: Plane of thermocouple leads at right angles to axis of F

first soldering the resistor, R, to the thermoelement, TE, and to the coaxial input connector, A, which is mounted in disk, Di. This disc is then fastened to the cylinder, F, and the two thermocouple

leads are soldered to the 2-pin output connector, B. Disk Di, which has a small

hole with a cross wire at G, is then fastened

to F and the free end of the heater of the thermoelement is soldered to the cross

wire. For clarity both thermocouple

leads are shown in the figure. However

the plane of the thermocouple leads and

the pins of the output connector are ac- tually at right angles to the axis of the

cylinder so that minimum emf is induced

in the output circuit. In use F must be

grounded through the input connector. The 100- and 200-volt converters contain two resistors and two cylindrical inner shields. Si and St, as shown in Fig.

2, to reduce the effect of the distributed capacitance from the resistors to the outer cylinder. This considerably ex- tends their frequency range. Shield St

is centered in by a narrow ring, and is fastened to F by three equally spaced

screws. Slots in Fmake axial adjustment

of ^2 possible. The end-plate of 5i is soldered to the input lead and to the re- sistor in assembly and this shield is fast- ened to Di by small insulators. The

assembly is otherwise similar to that de- scribed for the other converters. The

50-volt converter has a single resistor and

a fixed inner shield like 5i in a cylinder of the dimensions given in the caption of

Fig. 1. Additional information on the voltage converters is given in the first three columns of Table I. The conmiercially

available deposited carbon resistors are 1/8 inch in diameter and 2 iaches long,

and are vacuum sealed within a 3/16-inchdiameter glass cylinder. Some resistors of a different construction showed unaccountable a-c-d-c errors above 10 mc

when used in earlier converters. The 5- and 10-ma uhf thermoelements, each with

colinear supports and heater wire, are

also commercially available, with an out- put voltage of about 7 mv (millivolts) at rated current. Each is in an acomshaped evacuated glass bulb, and has a

small bead between the heater and hot

junction of the thermocouple to instilate them electrically but not thermally.

A Lindeck potentiometer rather than a

millivoltmeter is connected to the output

emf of the converter to provide high

resolution. A circuit diagram is shown in Fig. 3. The potentiometer has ranges

of 0.15, 0.75, 7.5, and 15 mv. The two

lowest ranges are used for sensitivity checks. With a galvanometer having a

-^vYwwywwwwYWww0.1 a 0.4ft 4 5n ba

soon

ion

Fig. 3. Circuit oF Lindeck potentiometer

M= 1.5-ma d-c milliammeter K = Thermofree key

E = Emf terminals, ANiOSL

Gi, Gj = Galvanometer terminals, ANIOSL

Note: All fixed resistors of manganin,

accuracy +0.2%

sensitivity of 200 mm (milHmeter)/;ia

(microampere) it provides a selected res- olution of either 2 or 10 /iv (microvolts) per mm deflection. The potentiometer is completely shielded, and a 2-conductor

shielded cable connects the output of the converter to the terminals marked E in the figiu-e. The advantages of this po- tentiometer are its low thermal emf (less than 1 fiv) and its freedom from drift (much less than 0.01%/minute under

ordinary laboratory conditions).

Theory

Because of the geometry, the resistor, R, in the cylinder of each low-voltage converter (without inner shields) may be

represented as a transmission line of

length, (, with a imiformly distributed

series impedance of z ohms per unit length

and a uniformly distributed shunt adTable I. Voltage Converters

Thermoelement

Rated Rated

Voltage, Current,

Volts Ma

Heater Series Resistance, Resistor, Ohms Kilohms

1.3.

5. 10. 20.

SO.

100. 200.

. 5.

. 5.

. 5.

. 5. .10.

.10.

.10. .10.

.90.

.90. .90.

.90.

.25. .25.

.25.

.25.

. 0.2

. 0.5

. 1

. 2

. 2

. 5 .10»

.20*

* Two equal resistors in conBguration of Fig. 2.

157
•004

•002

0 0

-0.02

-0.04

-0.06

-OOB

-010

-0.12

-0 14

K-0

038 -y^

K=0.03

r i 1 1 )

1

K.-O.Z K=0.3

K-0.4

Fig. 4. Computed «-c-d-c

difference at a function of tlie parameters a and

k

mittance of y mhos per unit length. The

line is terminated by the heater of the thermoelement. We are interested in the magnitude of the transimpedance Zc = Vi/Io where F< is the input voltage

being measured and /<, is the current through the heater. If the heater resist- ance, Rh, is much less than that of the

resistor, R, we may to a first approximation consider this a short-circxiited line. It will be apparent later that this is not

a drastic approximation at low frequencies

since we are concerned only with the effect of the line reactance on Zc- Then by

ordinary steady-state transmission-line

theory,

sinh y/Z Y (1)

where Z = zf=R+jo}L, and V=yi=jo)C

and R, L, and C are the total resistance,

series inductance, and shunt capacitance

of the line. We are interested in the low- frequency range in which the effects of the reactances are small. We may define the parameters a = uCR

and b=wL/R. If each of these is less than unity the hyperbolic function may

be approximated by the first three terms

of its series expansion, so that ZY {ZYY

Then if all terms in a^ft"" for which «i+

w>3 are discarded we find that (ab a' ja\

and, to the same order,

'

' \ 90 3 /

/ a» 6' ab\ , , i?H 1 I (2) \ 180 2 6 /

The a-c-d-c difference of a voltage converter is defined as

Vac- Vdc

Vic

(3)

where Vac and V^c are the rms alternating

and direct voltages required to obtain the same response (output emf) of the converter. If the thermoelement has no

a-c-d-c difference and if we define k = h/a,

this becomes

So = -\zAR

where So signifies that other possible causes of a-c-d-c differences are neglected.

The relation makes it possible to esti- mate So to this approximation very simply. For the cylindrical construction of

Fig. 1 with a cylindrical unspiraled resistor, L = 0.012^M microhenry and

C= 0. 61 C/M picofarad, where M=

logio(g/A) and g and h are the diameters

of the cylinder and resistor, respectively,

and e is in inches. Thus niunerically,

k=l.9X10*iM/Ry and a = 3.8X10-"

flR/M, where / is the frequency.

For example if i? = 2,000 ohms, g and h

are 2 in (inches) and 1/8 in., and ^=2.0

in., 5„= +0.001 (+0.1%) at 40 mc.

It is apparent from equation 4 that to

this approximation certain values of k should make So = 0. These are ki = 0.038 and ^2 = 0.30. The mmierical values of re- sistance required are Ri = 710Af and Ri =

25GM. Unfortunately the logarithmic

relationship for M permits little range in R for reasonable values of g and h, for if g is greater than the length of the resistor, end effects may become pronounced.

It is of interest to extend the analysis

to higher values of the parameter, o. This has been done by solving equation 1 for Zc with the aid of Kennelley's tables and graphs of hyperbolic functions having

complex arguments,^ and then computing So = {\Zt\ —R)/R. The results are shown in Fig. 4. It is rather striking that

the first solution of equation 4, ki = 0.038,

results in reasonably small a-c-d-c differ- neces (<5%) to a = 10, a range much

greater than that permitted by the initial assumptions. For low-voltage converters the effect of the heater resistance R/, cannot be neglected. To the next approximation the

heater can be considered as a lumped

resistance termination for the trans- mission line. Then from steady-state

transmission-line equations.

Zc--Rh cosh VZY+ sinhVZF (5)

By expanding the hj^jerbolic fvmctions and discarding higher order terms as before (for a<l and &<1) we find that if R„<R

So (iT-J(---f)

-Rt_l( a ~2

(6)

where R, = R+Rh, Rh/R=m, A=m*+

2m/3+l/3 and B = m^+2m/5+l/15.

Table II. Comparative A-C-D-C Difference* of Voltage Converters

Voltage Range

Comparative A-C-D-C Difference, D, Per Cent

nverter Converter Applied A B Volts 0.1 mc 1 mc 5 mc 10 mc 20 mc 30 mc 40 mc

1 , .

. 3 1 . -1-0.01. . -0,08

1 3 ... 1.5... . -1-0.01.

.

. -0 01.

.

. 0.00.. . -0.02.. . -0.03 . . -0.0«

3 5 ... 2 .. -0.06

3 5 ... 3 .. . 0.00.. . 0.00.. , 0.00. . -0.01. . . -0.01.

.

.-0.03 . . -0.04

5 . ... 10 ... 3 .. +0.14

5 10 . . . 5 .

.

. 0,00.. . 0.00.. . -0.01. . -1-0.01.

.

. -1-0.02.. .+0.07 . .+0.16

10 .... 20 ... 6 .. -0.02

10 20 ...10 .. . 0.00.. 0.00. . 0.00.. .-0.01 . . -0.02

20 50.... 15 .

.

. 0.00. -0.04

20 60.... 0.00. . 0,00, .-0.01 . . -0.04

20 100 ...20 .. . 0.00.. . 0.00.. . +0.02.

.

, +0.07. . , +0,24.

.

. +0.33*. .+0.17

50 ... 100 ...50 .. +0.33

50 200 ...60 .. .-1-0.01.. . 0.00.. . -0.02. . -0.06. . . -0.04.

.

.+0,19*. .+0.67

100 200 ...90 .. . 0.00,. -0,14

Note: D = 100(Sb — Sa), where S is defined by equation 3, and the subscripts refer to converters A and B.

* Additional comparisons made at 25 and 35 mc.

158
The values of S„ for the 1- to 20-volt converters have been calculated by equation 6 at a frequency of 40 mc, at which

a and b are still less than unity. The

results are given in the last column of Table II. When )fe<0.038, distributed capacitance

from the resistor to the cylinder pre- dominates. At a given frequency and

geometry, the resultant error is pro- portional to the square of the resistance.

It can be minimized by the construction shown in Fig. 2 in which two inner cylindrical shields are used, with one con- nected to the input lead. Thus there are now two transmission lines in series. If Ri,«R, application of transmission line equations, with careful attention to

signs, gives for the desired transimpedance,

2c= -^-^^ tanh VzTk; cosh \/zryi+

^%inh v/zTf; (7)

where the subscripts 1 and 2 refer to the

left and right hand lines (resistors and

shields) of Fig. 2, and the resistance of the thermoelement is neglected.

If Zi=Z2 = Zand F,= F2= Fwehave

2Z

Thus the transimpedance is twice that

of either line, and the a-c-d-c difference.

So, will be that of half the total line. The

parameter, a, of equation 4 will then be

one-fourth that for a line of the same

dimensions without inner shields. Thus

at a given frequency for which a<l the use of the double shield rather than a sin- gle cylinder of the same diameter should

reduce the frequency error, for resistors of the same value and dimensions, by a

factor of 16. However, more detailed analysis in- dicates that even further improvement

may be possible. For these converters fe<<0.038, so that the series inductance

oi each line may be neglected. By ex- panding the hyperbolic functions of equation 7 and discarding higher order terms

it can be shown by lengthy and rather

tedious algebraic manipulation that So~0

when = 1.032, where d and (2 are the

lengths of the left and right hand lines of

Fig. 2, anda<I.

For the convenient dimensions shown in Fig. 1, the parameter, a, of each lowvoltage (up through 20 volts) converter now being used is low enough that it has

not been necessary to proportion the converters for minimum error in accordance

with equation 6. The higher voltage elements are constructed with dual shields, with the grounded shield axially adjustable. This shield is set to give mininnini frequency error as explained in the next

section. A useful theoretical study of coaxial

lines, terminated by resistors, was published by Crosby and Pennypacker.'

They show conditions for minimizing the input reactance.

Tests

Although long-time stability is not

required of these converters, fluctuations or drifts in emf for the short time between

the a-c use and the d-c calibration must

be less than the desired accuracy. Such

changes can arise from self-heating effects and ambient temperature changes, and

from thermal emfs and other changes in the Lindeck potentiometer. Tests have

shown that the self-heating of the carbon

resistors, which have a load coefficient of about 1% per watt, is the largest source

of drift. The change in resistance is very

nearly exponential, with a time-constant

of about 2 minutes. Since d-c calibra- tions can easily be made within 30 seconds

of the a-c readings, this self-heating error

is not significant in a-c tests if a short warm-up period is allowed. It is almost

completely eliminated by the procedure

used for a-c—d-c transfer tests. The effect of ambient temperature changes—the thermoelements have temperature coefficients up to 0.1%/C (degrees centi- grade)—should also be insignificant in a

laboratory with reasonable temperature

control. A great advantage of thermoelements

for a-c-d-c transfer measurements is the almost complete electrical isolation of the input and output circuits. The uhf thermoelements have a small electrically

insulating bestd between the heater and

hot junction of the thermocouple to eliminate conductive coupling. To mini- mize mutual inductance the plane of the thermocouple leads is at right angles to the heater and its colinear supports. Tests showed that at 4 mc the induced

a-c voltage in the output circuit of each

thermoelement was less than 20 fiv, the

resolution of the detector used. Induced

currents in the thermocouple circuit can

cause errors by joule heating of the thermocouple, but calculations indicate

this should not be significant if the induced

voltage is less than a few millivolts. The shield of the Lindeck potentiometer

provides reasonable immunity from in- duced fields. In some tests at 10 mc a

current of 2 ma from the shield to ground

through the lead from the potentiometer

to the converter caused no significant

error. For transfer meastircnicnts tiic nios( important requirement is that the a-c-d c difference of each of these converters be known to the full accuracy desired. The

general principles on which such deter- minations are based have been given.' In the frequency range studied, the major

error of each of these converters is caused by the reactance. For a given converter

this error should be independent of volt- age level. Thus, it was feasible to eval- uate the relative errors by intercomparing converters of adjacent voltage ranges

to determine their differences in fre- quency response. A complete series of such comparisons was made at two volt- age levels for each pair of converters, at frequencies up to 40 mc. In each comparison the two converters were connected

in parallel to a coaxial lead through a tee

fitting {GR 874), and a shielded potenti- ometer was connected to each output.

Each potentiometer was adjusted for zero deflection at the test voltage. The

converters were then supplied in succession with alternating, direct, reverseddirect, and alternating voltage. Each

voltage was adjusted to produce the same emf of the higher range converter and the

deflection of the galvanometer connected

to the other converter was observed. From the differences in emf, directly

determined from these differences in deflection, the difference in the frequency

response of the two converters was deter- mined.' In these tests the movable inner

shields of the 100- and 200-volt converters were adjusted for best performance over

the desired frequency range, by comparison with the 20- and 50-volt converters. The errors of these adjustable converters were found to be complicated functions of the frequency and shield position.

The results of these intercomparisons

from 0.1 to 40 mc are given in Table II. They show that for each pair of converters

the relative a-c-d-c differences were in- dependent of the applied voltage to 0.02% or less, and were less than 0.4% up

to 30 mc. For the low-voltage converters they were less than 0.05% to 20 mc

without exception and without evidence

of systematic errors. The relative a-c-d-c differences of most

of the thermoelements used in the voltage

converters were also determined before

the thermoelements were installed, by

making similar a-c-d-c intercomparisons

as ciurent-measuring elements at 40 mc.

For these tests the two thermoelements

were connected in series along the axis of a brass cylinder having the same dimen159
Table III. Results of Test of 1-Volt Voltmeter

Frequency, Applied Voltage,* Mc Volts

1 0.997

10 0.998

30 0.998

100 0.999

200 0.982

300 0.945

400 0.883

*For same reading of millivoltmeter at each fre- quency

sions as in Fig. 1 but with a coaxial input connector at each end and with two 2-pin

output connectors. A split cylinder with one part readily removable facilitated changing thermoelements. The input was applied to one connector with the

other short-circuited. Rough calcula- tions indicated that at 40 mc the current should change by about 0.5% along the transmission line formed by the heaters of the thermoelements in this cylinder, but

that the distribution should not be greatly

dependent on the resistance of the heaters,

less than 100 ohms each. Therefore, two

determinations of relative a-c-d-c differ- ences were made with each pair of thermoelements, with first one end then the other

of the cylinder short-circuited. For each

pair the two determinations differed by

about 0,4% at 40 mc, but in each case

their algebraic average was less than

0.02%.

These calculations and tests also provide assurance that even at 40 mc

the ciurent along the heater of a single

thermoelement terminating a voltage converter is well within 0.1% of the value

at the mid-point of the heater. The 1-volt converter was compared at rated voltage with a wire-wound thermal

voltage converter of known a-c-d-c differ- ence at 3 cps and 20 kc, with observed

differences of +0.02% and 0.00% respectively.

The response of the 1-volt converter, with a shielded d-c miUivoltmeter con- nected to its output, was determined to better than 1% at frequencies from 1 to 400 mc by the bolometer bridge of Selby

and Behrent,* with the results shown in Table III. It is evident that the fre- quency influence is very small to 100 mc.

It increases rapidly at higher frequencies,

becoming —5.2% at 300 mc. The cal- culated a-c-d-c difference at 300 mc by

equation 6 is only —0.9%; this converter has a 1/4-in. -diameter unsealed resistor. Some of the discrepancy may be accounted

for by the voltage rise in the connector. A type N-UG58/U input connector was

substituted for the type 574 for these

tests.

As a check on equation 6 at larger a-cd-c differences, the 50-volt converter was

tested before its inner shield was installed. The measured a-c-d-c difference at 40 mc was -j-1,3% and was accurately pro- portional to the square of the frequency.

The computed values were 40% smaller, indicating that the effective length of the

resistor, the end effect, was 1,2 times the actual length. At 40 mc the measured a-c-d-c difference was changed

by less than 0,02% when the resistor was mounted 3/32 inch off the axis of the cylinder, indicating that exact center- ing is not critical even when the

errors are large. The intercomparisons of Table II show

that the a-c-d-c differences of the five low-voltage converters without inner

shields, and with 1- to 20-volt ranges, are

all equal to better than 0,05% to more

than 20 mc. At 40 mc they agree to 0,1%

with the values calculated by equation 6. Values cannot readily be computed for the higher voltage elements with the inner

shields. This imanimity between converters having such a wide range of re- sistors and different thermoelements gives

good assurance against unknown sources

of error. It is quite unlikely, but not

impossible, that each converter would

have the same a-c-d-c error. However

the test of the 1-volt converter with the bolometer bridge provides most valuable

additional assurance. The large errors of this converter above 100 mc decrease

rapidly with decreasing frequency, well within the stated accuracy of the bridge measurements. For converters without

inner shields almost all known causes of such errors, such as the effect of reactance, equation 6, skin effects in the resistor or thermoelement, etc, should cause a-c-d-c

differences approximately proportional to the square of the frequency, over the range for which the errors are small. Thus the authors beheve that the large

measured errors above 100 mc can be

extrapolated downward to lower fre- quencies to indicate with considerable confidence that the a-c-d-c difference of

this converter is less than 0,2% at 40 mc

and less than 0,1% below 30 mc. Based on the foregoing consideration the authors assigned a value of zero to the

a-c-d-c difference of the 1-volt converter

to 40 mc, and then determined the a-cd-c differences of all the other converters from the intercomparison data in Table

II, The results, rounded to the nearest 0.05% to 10 mc and 0.1% to 40 mc, are given as observed values in Table IV.

Use

These converters are most conveniently

used to meastire the a-c-d-c difference or frequency influence of other rms instru- ments such as thermocouple voltmeters,

which are now available with accuracies of 1/2% to 10 mc, A typical setup for this

is shown in Fig, 5. The scale calibration

of the voltmeter can then easily be

checked on reversed direct current. The

a-c-d-c difference tests are similar to the intercomparisons already described, and

d-c calibration of the converter is not

necessary once the scale factor of the Lindeck potentiometer is determined for each converter. This is the per-cent

change in input voltage per centimeter

change of galvanometer deflection.* Either direct or low-frequency alternating current may be used as the reference fre- quency and the test may be made rapidly

and acctirately. The results are only

sUghtly affected by drifts in either instru- ment. At frequencies above about 20 mc a small lead correction may be neces- sary if the connectors between the junction plane and the two instruments are not electrically equal, but this is readily

determined to the required accuracy. For a-c measurements a d-c potenti- ometer of 0.1% acciiracy or better is re- quired to measure the d-c reference volt- age. A deflection potentiometer and volt box, or an automatic self-balancing po- tentiometer ("digital voltmeter") should be convenient for this. For testing elec- tronic voltmeters frequency -response measurements (differences in reading for

Table IV. A-C-D-C Differences of Voltage Converters

Per-Cent A-C-D-C Differencee, lOOS

Rated Observed Calculated

Voltage, —

Volts to 5 mc 10 mc 20 mc 30 mc 40 mc 40 mc

1 <0.05 <0.05 0.0 0.0 0.0 0.0

3 <0.05 <0.05 0.0 0.0 -0.1 0.0

5 <0.05 <0.05 0.0 -0.1 -0.1 0.0

10 <0.05 <0.05 0.0 0.0 0,0 -FO.l 20 <0.05 <0.05 0.0 0.0 0.0 -fO.l 50 <0.05 <0.05 0.0 0.0 0.0

100 <0.05 -t-0.05 +0.2 -hO.3 +0,1

200 <0.05 -0,05 0,0 +0.2 +0.5

160
Fig. 5. 5-volt converter letup for a-cd-c test of experi- mental thermo- couple voltmeter. Lindeck potenti- ometer at right, double-shield converter and experi- mental split-shield element in fore- ground, a-c-d-c

switch not shown

the same voltage at the test and reference

frequencies^ should be particularly con- venient The form factor and crest fac- tor of the a-c sources must be within 0. 1%

of the values for a sine wave, 1.111 and

1.414 respectively, since most electronic voltmeters respond essentially to the aver- age or crest (peak) values. Since these

factors depend upon the phase angle as

well as the magnitude of each harmonic it would ordinarily be necessary to make

sure that the ratio of the magnitude of the nth harmonic to the fundamental does

not exceed 0.1xn% when an averagereading instnunent is tested and 0.1%

when a crest-reading instrument is tested. The results should then differ from those obtained with a sine wave by less than

0.1%. An r-f generator of at least 5 watts out- put and good voltage stability is required

but only moderate frequency accuracy and

stability are needed. It is very diffi- cult to construct broad-band generators

of good wave form at this power level. However, a simple adjustable L/C (in- ductance capacitance) tuned circuit, with

the instruments connected across the

capacitor, can be used simultaneously to improve the wave form, match the impedance of the instruments to that of usual 50-ohm source, eliminate the capacitance

Discussion

Morley J. Lush (Rawson Electrical Instru- ment Co., Cambridge, Mass.): I feel that

the instruments described in this paper are a real contribution to the art of accurate

voltage measurements at high frequencies.

Although it is difiRcuIt to make resistors with low reactances, Mr. Hennach has shown

that these can be obtained, and that volt- meters can be constructed which have a

flat frequency response all the way from

direct current out to the megacycle region.

The frequency response will remain stable

loading of the instruments on the source, and provide the higher voltages often re- quired. For these combined purposes

the values of inductance and capacitance should be chosen so that <aL=l/c^C^

iRi, where Ri and Ri are respectively the resistances of the soiu^ce, including the

inductor, and the load (instruments). For low voltages a 50-ohm resistor can be connected in series with the L/C cir- cuit and the instruments and attenuators

as needed can be connected in parallel

with this resistor for improved wave

form.

Conclusions

Cylindrical film resistors in series with a thermoelement in a coaxial line makes

possible single-range thermal voltage converters that are useful as a-c-d-c

transfer instruments over a very wide

frequency range, 3 cps to at least 30 mc,

with unusually high accuracy. They are inexpensive and easy to construct. Their

frequency influence may be estimated by

reasonably simple equations, with results which agree well with the measured values up to 40 mc. The voltage converters may

be useful to considerably higher fre- quencies. The higher range converters

require considerable power, up to 2 watts,

for long periods of time, so only d-c calibra- tions are needed, once the frequency

response has been determined. I have

found from my own work that accuracies of 1% can be obtained at frequencies up to 100 mc and now these instruments are available on the market. A precision of 0.1% can be obtained, but the absolute accuracy is limited by the lack of a primary

standard. I have been told that the National Bureau of Standards expects to be

able to offer 0.25% certifications by the end

of 1960. Other workers in this field have used a

fixed or variable capacitor as the attenuator

and have a marked but short warm-up

drift. These disadvantages could be re- duced by using metal-film resistors of higher resistance with a 5-ma thermoelement, but this would very probably re- duce the frequency range for the same

attainable accuracy. The good perform- ance of these converters was obtained with

deposited-carbon resistors of a new type.

Earlier converters with resistors of differ- ent construction showed discrepancies up

to 0.5% at 40 mc. Further work is planned to determine the cause of these

differences. These voltage converters can be used

quickly and easily to make a c-d-c differ- ence tests to determine the frequency in- fluence of other rms instruments to 0.1%

or better to at least 10 mc and to 0.2%

at 30 mc. Direct a-c measurements are made by the transfer technique, which

in most cases can be arranged for reasonable simplicity. With soiu-ces of suit- able waveform, average-reading and crest reading instnmients could also be cali- brated if desired. In all of these applications the applied frequency need not be

closely determined or held because of the

flat frequency response of these voltage

converters.

References

1. Electrical Indicating Instruments. ASA

€39,1-1959, American Standards AssociBtioo, Inc., New York, N. V., sect. 3.11.3. May 1959.

2. Thermal Converters as AC-DC Transfer

Standards for Current and Voltage Msasursments at Audio Frbquencibs, F. L. Hermach.

Journal of Research, National Bureau of Standards, Washington, D. C, vol. 48. 1952, pp. 121-38.

3. A Wide-Range Volt-Ampere Converter

FOR Current and Voltage Measurements,

F. L. Hermach, E. S. Williami. AIEE Transactions, pt I {Communication and Electronics) , vol. 78, Sept. 1959, pp. 384-88.

4. Tables and Chart Atlas of Complex

Hyperbolic and Circular Functions, A. E. Eennelly. Harvard University Press. Cambridge,

Mass., 1921.

5. Radio-Frequency Resistors as Uniform

Transmission Lines, D. R. Crosby, C. H. Pennypacker. Proceedints. Institute of Radio Engineers, New York, N. Y., vol. 34, 1946. pp. 62-66.

6. A Bolometer Bridge for St.\ndardizino

Radio-Frequency Voltmeters, M. C. Selby,

L. F. Behrent. Journal of Research, National Bureau of Standards, vol. 44, 1950, pp. 15-30.

element in series with the thermocouple. It

is possible to make an excellent low-loss capacitor using a micrometer screw for ad- justment. A wide frequency range and

wide voltage range can be obtained on one

instrument. However, the voltmeter can- not then be used for d-c to a< transfer measurements, since it has lero response at zero frequency. Indeed, the response is directly proportional to the frequency, and

it is necessary to calibrate the voltmeter at each voltage and frequency for which it will be used.

It should be pointed out that connections

to these voltmeters are a problem, as they

161
are in all higb-frequency apparatus. The

use of coaxial coonectors can be misleading,

as any appreciable length of 50-ohm coaxial

line will cause errors due to standing waves,

unless the voltmeter input impedance is made exactly equal to the impedance of the

line. Careful matching of the line impedance will certainly be necessary if volt- meters of this type are to be used at fre- quencies above 100 mc.

F. L. Hermach and £. S. Williams: Mr.

Lush's remarks are appreciated. We believe these thermal voltage converters, by

taking advantage of the powerful a-c-d-c

transfer technique, should make possible a new order of accuracy in a-c voltage measurements up to at least 30 mc. Their

usefulness will probably fall off rapidly at higher frequencies, except for the lower volt- age ranges, up to perhaps 10 volts. A major advantage of an a-c-d< transfer standard is that its a-c-d-c difference is relatively permanent so that once evaluated,

it should not ordinarily need to be redeter- mined. Thus, elaborate repeated a-c cali- brations are not required. This and the

ease and simplicity of constructing these converters should offset the extra labor of making the a-c to d-c transfer for each measurement. We mentioned connection errors only

briefly. Each converter is a high-iinped- ance load at tlie end of a short coaxial line from the junction plane to the converter.

If this line is a small fraction of a wavelength

the voltage rise from the junction to the end

of the line is approximately 50A'%, where h^uCZo and C and Zo are the total shunt

capacitance and the characteristic impedance of the line. At 30 mc for a 2-inch length of typical 50-ohm line the correction

is only 0.1%. It is offset by a similar correction if an equal line is used between

the junction plane and the test instrument or load to be measured. .\t higher frequencies

or for longer lines the correction can be

large and this approximate formula may

no longer be valid. In such cases, however, voltage nieasureiiicnts probably have meaning only witli lines terminated in their characteristic impedance, so that the standing-wave ratio is close to unity.

162
Section 11

Journal of Research of the National Bureau of Standards Vol. 48, No. 2, February 1952 Research Paper 2296

Thermal Converters as AC-DC Transfer Standards for

Current and Voltage Measurements at Audio Frequencies

Francis L. Hennach

Thermal converters and associated equipment that are used as ac-dc transfer standards

at the National Bureau of Standards for the precise measurement of current and voltage at power and audio frequencies are described. The standards and the equipment are primarily

used to standardize a-c ammeters and voltmeters submitted to the Bureau for certification. The ac-dc transfer may be made with these thermal converters at currents from 1 milliampere

to 50 amperes, voltages of 0.2 to 750 volts, with an accuracy of 0.01 percent at frequencies

from 25 to 20,000 cycles per second. The special tests to insure the required accuracy of the transfer standards are described,

and the results are presented. A number of factors that limit the transfer accuracy of ther- mal converters have been discovered, and the results of special tests and theoretical work to evaluate these factors are discussed. The solutions, by an approximation method, of certain pertinent nonlinear differential equations governing the heating of a conductor by an electric current are given.

1. Introduction

The increasing use of electric energy for aircraft,

induction furnaces, and induction heating, and the

greater accuracy required in measurements in electronics, have led to increasing demands for the

accurate standardization of ammeters and voltmeters at frequencies extending upward from power

frequencies through the entire audio-frequency

range. To meet these demands, special instruments

have been developed at the National Bureau of

Standards for the measurement of current and

voltage over rather wide ranges. They make use

of thermal converters ' (often called thermoelements)

like those incorporated in ordinary thermocouple

instruments, but differ in the manner of reading

and use. They may be used either directly to measure the ac-dc differences of ammeters and

voltmeters, or with a suitable potentiometer and

accessories to measure alternating currents and

voltages. They were designed and are used primarily for testing electric instruments, at currents

from 1 ma to 50 amp and voltages from 0.2 to 750 v, with an accuracy of 0.01 percent at frequencies

from 25 to 20,000 c/s.

2. Transfer Principle

The basic electrical units are defined in terms of a concordant system of mechanical units and are

realized by absolute electrical measurements carried

out at national standardizing laboratories to fix the value of groups of standard cells and resistors.

These standards are used in conjunction with a

potentiometer to make measurements of direct

voltage, current, and power. It has been known

for a long time, but not sufficiently realized, that

the measurement of the corresponding alternating-

' Proposed AIEE definition 30.89.040: A thermal converter is a device that

consists of one or more thermojunctions in thermal contact with an electric heater or integral therewith, so that the electromotive force developed at Its output terminals by thermoelectric action gives a measure of the input current

in its heater.

205001-52 1

current quantities depends fundamentally on certain

standard types of electric transfer instruments that

ideally have the same response on direct and alter- nating current. If the instruments are calibrated on direct current at the time of each use and if the

precision of reading is suitably increased, long-time

stability, freedom from drift, small temperature

influence, low losses, and other normally desirable

characteristics become of secondary importance,

and the instruments may be primarily designed and

constructed to have the best possible frequency

characteristic. For difference measurements such

instruments need not have long scales in the ordinary

sense, but rather may be designed for increased

precision of reading by methods that would not be

applicable to instruments intended for general

service.

Special electrodynamic transfer instruments have

long been used at the Bureau for accurate a-c measurements. These transfer standards are used

largely to standardize (test) other instruments

submitted to the Bureau for certification. There

are two distinct types of such tests. For the

"straight a-c test", the instrument under test and

a transfer standard are connected to measure the same alternating electrical quantity (current, volt- age, or power), which is adjusted to produce the

desired deflection of the test instrument. The

response of the standard instrument is observed,

then the standard is transferred to direct current. The direct quantity is adjusted to give the same

response of the standard and is then measured with

a suitable potentiometer and accessories. For the

so-called "ac-dc difference test", both instruments

are connected to measure the same quantity first on alternating and then on direct current, which

is in each case adjusted to give the same deflection

of the test instrument. From the averaged difference

in the response of the transfer standard, the ac-dc

difference of the test instrument is computed.

The second type of test can be made with somewhat

higher accuracy than the first, and in conjunction

163
with a test on direct current gives more information

about the performance of the test instrument.

Because the ac-dc difference of an instrument depends upon geometrical factors that are relatively

permanent, subsequent tests for checking the con- stancy of calibration need generally be made only on direct current. Occasionally transfer tests are made by using a selected low frequency (such as 60 c/s) as the reference in place of direct current. An example of the accuracy with which such

tests are made, laboratory standard instruments,

with scales 12 in. long, are regularly standarjdized

and certified at the Bureau to 0.05 of a scale division.

For a 150-division instrument this requires an accuracy of measurement of 0.03 percent or better.

3. Development and Description of Electrothermic Transfer Standards

3.1. Choice of Standards

Several types of instruments were considered in

choosing transfer standards to cover the full audiofrequency range. Electrodynamic instruments have

been highly developed [1, 2, 3]^ at the Bureau for

measurements at the commercially important power

frequencies and can be used with proper corrections,

up to about 2,000 c/s. However, unavoidable inductance and stray capacitance errors limit their

useful frequency range and present little hope for the

desired extension. Electrostatic instruments have

been carefully studied [4, 5] at the National Physical

Laboratory. They are useful over wide frequency

ranges, but because of their low-torque-weight ratio

at low voltage are not readily adaptable for either

the low voltage or the current ranges desired in this

application. Electro thermic instruments, which use an effect produced by the heating of a conductor

carrying a current to be measured, seemed more

promising. Of the several kinds, thermocouple in- struments, in which the temperature rise of the con- ductor (heater) is measured by a thermocouple,

seemed most feasible. Straight-wire heaters allow a wide frequency range, and the thermocouple

measurements permit high precision of reading. The

usual disadvantages of poor stability and large temperature influence are not important in their use as transfer instruments, and their low overload capacity is not a serious limitation in careful laboratory

work. Fortunately, good thermal converters are commercially available. They have been used for

measurements at frequencies up to about 65 Mc, with a d-c millivoltmeter as the indicator, and are generally considered in the Yi- to 2-percent accuracy

classes. So far as is known, they have not been

previously studied for use as transfer instruments at the frequencies and the accuracies needed in this

application.

For these reasons, commercial thermal converters were selected for study to determine their value as ac-dc standards in a wide-range audio-frequency

> Figures in brackets indicate the literature references at the end of this paper.

transfer voltmeter. A careful experimental and

theoretical study of their performance led to the

purchase of additional commercial thermal converters

of suitable ranges and the design and construction of

equipment for using them as transfer standards for

voltage and current.

3.2. Description of Transfer Standards

Each thermal converter purchased in the milli- ampere ranges is of the vacuum type^ mounted in an

evacuated glass bulb with supports of copper wire

embedded in the glass. The conductor, heated by

the current to be measured, is a short straight wire

generally less than Yi cm long and oft^en less than

0.001 in. in diameter. The heater alloy and dimensions are chosen to give a temperature rise of about

200° C at the center where the hot junction of the

thermocouple is fastened by a ceramic bead. This

bead provides electrical but not thermal insulation

between the heater and thermocouple circuits, with a

coupling capacitance less than 1 nni and an insulation

resistance greater than 50 megohms at 25° C.

Because high current sensitivity is not required,

the thermal converters in the 1- to 50-amp ranges are not evacuated. The hot junction of the thermocouple is welded directly to the midpoint of a thinwalled tubular heater. These thermal converters

are temperature compensated [6]. For current measurements, thermal converters in

the series 1, 2, 5, 10, etc., with rated output electromotive forces of 10 mv were purchased, with ranges

from 1 ma to 50 amp. Thermal converters of 7.5- and 30-ma ratings, with appropriate series resistors, are used for voltage measurements. Figures 1 and 2

show some of the thermal converters and the model

A and model B voltmeter elements. It should be

emphasized that the caption NBS on the nameplate

shown in figure 2 signifies only that the equipment was assembled for use at the National Bureau of

Standards. All the thermal converters were purchased commercially, as were the components of the

voltmeters. It should also be emphasized that the

thermal converters shown in these figures are the

transfer elements only. An indicator (to be discussed in the next section) is necessary for all measurements, and a potentiometer and accessories are

also necessary, except for transfer tests of otlier

instruments.

The model A voltmeter was developed as a prototype instrument to meet an immediate need for

voltage measurements. It consists of a 30-ma

thermal converter connected with its heater in

series with a commercial decade resistance box,

modified to have two 1 ,000-ohm-per-step decades,

and a 100-, a 10- and a 1-ohm-per-step decade, all in series. As shown in figure 3, one end of the heater

of the thermal converter is connected to the low-side

terminal of the box, with the shield of the box con- nected to the other end of the heater and to the

shield of the coaxial cable connecting the instrument

to the circuit. The center lead of tliis cable is con- nected directly to the high-side terminal of the box.

164
Fini KH ]. 7\i/pic(il iheriiial convciicrs.

FlOURE 2. Mounted thennal cojiverlers and the model A and

model B voltmeter elements.

Like the elcctiodynamic tiansfor standards used at lower frequencies, the series resistor is adjusted in use to give the same nominal current level for each

measured voltage. This was chosen at 20 ma, re- sulting in a voltmeter having a constant 50 ohms/v

and an upper range of 400 v. The model A voltmeter, described in some detail in an earlier paper

[7] is now no longer used for routine measurements.

For the model B voltmeter, a built-in 7.5-ma

thermal converter was used, with a fixed resistor

having taps to give voltage ranges of 1.5, 3, 6, 7.5,

15, 30, 60, 75, 150, 300, 600, and 750v. The resist- ance cards for this instrument were purchased comTnerciall}'^, and the cards and thermal converter were

inountcd in a suitable shicildcd box uitli the sludrl

connected to the low-side terrninul of the instruirHtiit, as shown in figure 3. For botli insti umcrits f lie cdects of capac^itance curicnts uiul of tiie self- inductance of the resistors were carefully considered;

the limiting factors arc discussed in another section

of this paper.

Shield

Thermol Converter

Shield Thermol Converter

790 600 300 150 75 60 30 15 7 5 6 3 1.5

FiauRE 3. Circuit diagrams of the model A and model B volt- meter elements.

A, Motlol A voltmeter olcmont; B, model B voltmeter element.

3.3. Indicator

Because of its convenience and portability, a milli- voltmeter is generally used for measuring the emf in the usual thermocouple instrument. The precision

of reading of the usual millivoltmeter is of course

far too low for this application. For increased ac- curacy a potentiometer can be used to measure the

thermocouple emf. However, if the thermal converter is used only as a transfer standard, the emf

need not be measured in a "straight a-c" test; and in ac-dc difference tests it is only the small change in emf

between the a-c and d-c settings that is significant.

The change and the full emf with wdiich it is compared need be measured with relatively low accuracy,

provided the change is referred to a highly stable

base value. Thus high sensitivity and high stability

but only moderate accuracy in the measurement of

emf are required of the indicator. For these i cquirements, a Lindeck potentiometer for providing the

base value, used in conjunction with a suitable galvanometer, the deflection of which indicates the

change, forms the ideal indicator. In a Lindeck

potentiometer, based on Poggendorf's second principle [8], the emf to be measured is balanced by an adjustment of the current through a fi.xed resistor.

At balance the voltage drop across the resistor,

which is the product of the potentiometer current

and the resistance, is equal to the measured emf.

The current is ordinarilv measured with a milliam165
meter, which generally sets the limit on the accuracy

obtainable with this form of potentiometer.

Such a potentiometer has been incorporated in a panel constructed for the audio-frequency testing. It was designed to have adequate stability, freedom from

changing extraneous electromotive force and the re- quired ranges, with sufficient precision of reading. Its circuit is shown in the central portion of figure 4. The resistor, R, is made of manganin, and a special

thermofree key [8] and copper binding posts are used

in the electrothermally sensitive emf circuit. The

components are mounted in the central part of the

panel, as shown in figure 5, and arc enclosed by a

3.4. Use of Instruments

These transfer standards arc used almost solely for

standardizing other a-c instruments. The necessary

switching and control circuits for such tests have been

incorporated in the panel shown in figure 5. In

addition to these circuits and the Ijindeck potentiometer, the panel contains the impedance-matching

transformer for the high-voltage circuits used in

testing voltmeters and the high-current circuits used

in testing ammeters. The present transformers provide ranges up to 50 amp and 800 v, with a nominal

input voltage of 150 v and a power level up to 100 w.

From AV

Supply From AC

Supply

AV

Tronsformers a Switch

AV input

From DV Supply

6 6

DV Input 6666

DC Input o o

AV-DV

Figure 4. Schematic diagram of circuits used for testing instruments. C signifies couxial lead with shield Rrounded; S signifles shielded lead with shield grounded.

grounded metal shield. No other thermal precautions were found necessary. A reflecting wallmounted galvanometer is used, with a voltage sensitivity (in the circuit) of 1 mm/nv. A No. 6 dry cell supplies the potentiometer current, which is adjustable from 0.2 to 1.2 ma. At these current levels,

repeated tests have shown that the steady drift in the

voltage across the four terminal resistor is less than

0.02 percent per hour, and that superposed fluctuations are much less than 0.01 percent. The fourterminal manganin resistor has taps at 0.1, 0.5, 2.5,

5, 10, and 25 ohms, giving milivolt ranges, at a

current of 1 ma, of these same numerical values.

The lowest range is used in transfer tests as a quick

check of the potentiometer. With the thermocouple

of the thermal converter connected to the "TC"

posts of the potentiometer, but with no current

through its heater, a galvanometer deflection of 10.0

cm to the left on the scale with the Lindeck set for 0.1 mv serves as a valuable partial check of the required

accuracy and correctness of the Lindeck circuit.

Motor-generator sets and voltage stabilizers are

available for tests at 60 c/s and audio-frequency

oscillators and power amplifiers for tests at other

frequencies. Batteries, controlled by suitable adjustable resistance voltage dividers and series resistors, are used for the necessary direct-current sources. The potentiometer, standard cells, volt boxes, and

resistors are those normally used for instrument

testing and are periodically standardized to insure a continued accuracy of considerably better than 0.01

percent.

A scliematic diagram of the major circuits and

equipment on this panel is shown in figure 4. For an

a-c test of an ammeter, the transfer thermoelement

is connected to the AC-DC binding posts and its thermocouple to the TC posts on the panel. The

test instrument is connected to the AC posts, and a

suitable four-terminal standard resistor is connected

to the DC posts for use in measuring the d-c current

by means of a potentiometer. For low currents at

tlic higher audio frequencies, shielded leads with the

166
shields grounded are used to minimize errors due to stray capacitance currents. The alternating current

is adjusted for the desired deflection of the test instrument, and the Lindeck potentiometer is adjusted for an "on-scale" deflection of the galvanometer. The

standard instrument is then switched to direct current, which is adjusted to give the same galvanometer

deflection and is then measured with the external

potentiometer. The direct current through the

heater of the tlicrmoelement is then reversed and the

d-c measurement repeated.^

Figure 5. I nsirumenl testing panel, including Lindeck

element.

For an a-c test of a voltmeter, a similar procedure

is used. The transfer standard is connected to the

AV-DV receptacle on the panel with a coaxial lead,

with due care to connect its thermal converter to the

grounded shield of this lead, and the thermocouple

is connected to the TC posts on the panel. The

test instrument is connected to the circuit with a

special twin lead consisting of two coaxial lines permanently paralleled at the instrument and connecting

to the two AV receptacles on the panel. Similar

leads of the same length are used to connect the high

side of a volt box for use with the external potentiometer to measure the direct voltage, to the two

DV receptacles. The use of these leads eliminates

the effect of any stray magnetic field or lead resistance on the measurements.

This simple procedure is reliable only if the drift

of the response of the standard instrument is negligibly small in the time taken for a series of such

t„15'"^'l '^^efsed d-c readings are common practice for highest accuracy with all types of ac-dc instruments.

readings. This is fortunately true for each of the

thermal converters used. After an initial warm-up

peiiod of a minute or so, the maximum obscrvfni

drift under those laboratory conditions of any of tiie thermal converters tested was less than 0.5 percent

per hour, and the average less than 0.05 percent per

hour. If it were necessary, the effect of significant

drift could be practically eliminated by using a deflection method as outlined for transfer tests, calibrating

the standard instrument on direct current at the

nominally correct cuncnt or voltage before and after

each a-c setting.

For ac-<lc difference or transfer tests, which are more frequent and usually more important than

straight a-c tests, the test and standard instruments

are connected in series to the AC-DC posts ^ in the

case of ammeters, or in parallel with suitable coaxial

leads to the AV-DV receptacles in the case of voltmeters. Suitable short leads are used to connect the

two AV receptacles together and to connect the two

DV receptacles together. The current or voltage is adjusted for the desired deflection of the test instrument, and the Lindeck potentiometer is adjusted for a deflection near the null position at the center of

the galvanometer scale. The two instruments are then in quick succession connected to alternating,

direct, reversed direct, and then alternating current. The cunent is adjusted to give the same deflection

of the test instrument in each case. The resulting

deflections of the galvanometer of the Lindeck potentiometer are observed with the potentiometer controls undisturbed. This procedure practically eliminates any effect of drift in either instrument, and

errors in repeating the settings of the instrument

under test can be made very small by using a lowpower microscope and setting the image of the

instrument pointer in coincidence with a cross-hair

in the eyepiece. The control circuits have been

arranged to facilitate switching, and with two

observers the procedure is rapid even with moderately fluctuating voltages as only one of the two

instruments is set to a given deflection.

From the average of the deflections on alternating

and on direct current, the ac-dc difference of the test instrument is readily computed by a simple formula.^

4. Evaluation of Transfer Performance

The ac-dc difference of each transfer standard

must be known to the full accuracy desired in the

measurements to be made. As, in a sense, there are no absolute transfer standards, such an evaluation

essentially consists of four steps.

L Choice of a type of instrument that theory indicates is suitable over the desired ranges.

2. Study of all known effects that can cause such an instrument to depart from this ideal.

3. Construction and use of an instrument in such

a way that these effects, by computation or

direct tests of the separate components, are

negligible or known.

< One terminal of the beater is connected to the i>ost that ta groanded on

alternating current.

' The formulas used in computing the results of the a-c and transfer tests are developed in appendix 1.

167
4. Comparison of actual transfer performance of

the standard with that of some other instrument, preferably of a C]\iite different type,

whose performance has also been evaluated.

The evaluation of these electrothermic transfer

standards was considered of ])rimar3'' importance

because of the high order of accuiacy demanded in this application. The unique facilities of the Bmeau

made this quite feasible, particidarly for step 4,

which is necessary to guard against eirors from un- suspected causes. Prototype instruments such as

these, in which previously available ranges are greatly extended, cannot be directly compared with

other types of standards over their full range. The

evaluation therefore consisted of the study, comp\itations, and special tests of steps 2 and 3 during the

design and constniction of these instruments and

their panel, the intercomparisons of thermal converters of adjacent ciurent ranges in the series, the

comparison at low frequencies of selected thermal

converters with the electrodynamic transfer standards at the Bureau, and the comparison of these

thermal converters with certain portable hot-wire

and electrostatic instruments also available.

A thermal converter can be used as a transfer

standard for alternating-ciUTent measurements over

the range of frequencies for which the response characteristic, E=J(i), is the same as with direct current,

where i^is the output emf, and / the current through

the heater of the thermal converter. For a voltmeter element consisting of a resistor in series with a thermal converter, the magnitude of the effective impedance Z=T7/ must also be equal to the effective

resistance on direct current, where V is the applied

voltage. In addition to these two requirements for

instrument testing, the fundamental principle that the

standard and the instrument under test "see" exactly

the same quantity must be carefully observed.

The response characteristic of the types of thermal

converters used were believed to be suitable at fre- quencies far higher than those used in this applicacation, and their direct-current and low-frequency

responses were verified by speci.-d tests described in other sections. The eflVctive impedance of each

voltmeter element was approximately comi)uted

from inductance and capacitance data suppli(>d by

the manufacturer of tlie resistance cards, and was

verified by special tests to be (](>sci-ibed later. The

capacitance and resistance of the Lindeck potiMitioineter to ground wei'e measured and found to be

1 ,()()() ii^i and 10, ()()() m(\g()luns at an i)ml)ient I'elative humidity of 40 percent. Thus the computed stray

currents are sufiiciently small so tliat no appreciable

fraction of the currents measured a\ ill l)e diverted

through the thermocouple at audio frequ(Micies when

one end of the heater is grounded. The use of

shielded leads when^ necessary, the avoidance of a ground between the test instrument and the standard

in the current circuits, and the use of the special coaxial lines in the voltage circuits assure that the

fundamental principle is observed even at the

smallest current and voltage in the present ranges.

The comparison tests of step 4 were planned to give maximum possible assurance against errors ihw

to unsuspected causes. Each comparison consisted

of repeated careful transfer (ac-dc difference) tests, as described in section 3.4. The 1-, 2- and 5-am])

thermal converters were compared with the NBS

electrodynamic transfer ammeter at selected fre- quencies from 25 to 200 c/s. The model A voltmeter was compared with the NBS suppressed-zero electrodynamic voltmeter at frequencies of 30 and 60 c/s and

voltages from 20 to 240 v. Both of these electrodynamic instruments have been carefully studied

[1, 2]. The\^ can be read with a precision of better

than 0.005 percent. The results of these comparisons, corrected for the small known transfer errors of

the electrodynamic instruments, are shown in table 1. They are estimated to be accurate to 0.005 percent or

better. The results of these tests disclosed that the

model A voltmeter and the electrodynamic voltmeter were in excellent agreement at these frequencies.

Table 1. Resulls of comparisons of Ihervial converters with other types of instruments

Thermal converter range

Model A vollineler- Do.... -.. Do

Do

Do

Do.. Do

other iiistrunieiit

Electrodyiiainic amineter. do

.do.

.do. .do.

.do. .do.

.do. Hot-wire animeter.

....do .

Test

current

or voltage

a Hi p

0.

Elect rod viiamic volliiieler do

do

do.. . .....do

Electrostal ie voltrnelrr. - do

M

12()

1.5() IW

3IK)

2,'> C/S %

-0.()#1 -.OKi

-.067 -.022

-. 11.5 -. \M

-.(KM

-. 010 -.Oil

30

e/s

-0.002

OIH

-(-. 00-1

(iO c/s %

-l-O. 001

"-.'oi:i -.(K)l -.02,5 -.(H.S

-.00(1

-.018

— . IHII — IMl.'i I

Comparative ac-dc dilTereiice

1(K) e/s

-0. (H)l -. OOS -.oos

001

-.(K)i» -.01,5

-.00.5

-.(H).5 -.(K)S

-AWl

-Am

21K) e/s

-0. mi

-. (K»i -.011.5

-l-.OIIl -t-.(Kll -.(KI2

1,000

c/s

10, (KH) e/s

+. OIH -.010

-0. (MH

1,5, (MM) e/s20. 000

e/s

-0. (M)i; -. (M)2

-. (MI2 (MW

168
They disclosod significant disagreements between the

iiinpeic-ninge tlicrmal converters and the electrodynamic ammeter, Avhich were considered to be due

to a low-frequency error in the 1- and 2-amp thermal

converters antl a small error independent of frequency

in the 5-amp thermal converter. The model A voltmeter was also compared with

two new high-grade portable pivoted electrostatic

voltmet(>rs, liaving scales over 5 in. long, at selected

frequencies from 30 to 20,000 c/s. For increased

precision a low-power microscope was focused on

the pointer of the electrostatic instrument under

test. This instrument was always set to the same

deflection and was tapped lightly before each reading, llepeated sets of careful readings were taken

at each frequency, and all tests were at the full-scale deflection of the electrostatic instrument. Because

of these repeated readings and the favorable conditions, the averaged results are estimated to be accurate to 0.01 percent. The results, as shown in table

1, verify the absence of capacitance errors in the

thermal voltmeter. Direct comparisons at the

higher frequencies could not be made at lower

voltages, but calculations indicate that skin effect and the effect of inductance should be considerably

less than 0.0 1 percent at 20,000 c/s. Similarly, the 5-amp thtrmal converter was compared with a nigh-quality portable 5-amp hot-wire

ammeter at frequencies from 25 to 20,000 c/s. The

results, as shown in table 1, indicate excellent agreement at frequencies of 1,000 c/s and above. Tests

at lower frequencies showed very puzzling discrepancies, which, in a separate investigation, were

traced to vibration erroi-s " in the hot-wire instrument. For that reason the results at frequencies

less than 1,000 c/s are not shown in this table. A plus sign in table 1 indicates that with equal

response of the thermal converter on direct and

alternating currents the response of the other instrument was greater on alternating than on direct current. Each thermal converter was compared with the

next higher- and lower-range thermal converter in

the series at two currents and at selected frequencies

from 25 to 20,000 c/s. The results of each of these

comparisons at the higher of the two currents at which tests were made are shown in table 2. They

are estimated to be accurate to somewhat better

than 0.01 percent, with very little possibility of

systematic error. A plus sign in the table indicates

that with equal response of the lower-range thermal

converter on alternating and on direct currents, the

response of the higher-range thermal converter was

greater on alternating than on direct currents. This

series of tests was made to extend stepwise the

results of low-frequency comparisons with the elec- trodynamic ammeter, and to indicate any highfrequency effects that might well be expected to

differ in different thermal converters.^ The group

• Electromaenetic forces between the hot wire and its cl.)sely adjacent iron mounting plate resuUod in vibration of the wire because the plate was masnetized by the damping niaenct of the instrument. This vibration caused additional

cooling and thus an error that was as large as 0.2 ;)era>nt at certain frequencies. 'The percentage errors due to any high-frequency effects would bo expected

to be independent of the current level and could thus be evaluated by these tests.

Table 2. Results of intercomparisons of thermal converters

of tests disclosed small discrepancies independent

of frequency in a few thermal converters, and verified the absence of any high-frequency errors in

the thermal converters tested. However, the}' could

not be used to assign numerical values to the errors

of individual thermal converters in the milhampcrc

range as the differences were found to depend upon

the current. The concluding series of tests to evaluate these errors were comparisons of a shunted

thermal converter with each of those pairs of thermal

converters that showed discrepancies in the previous

intercomparisons. The model A voltmeter was

shunted with a bifilar resistor of computed skin effect and time constant for tests at currents from 1 to 40

amps, and with a suitable high-quality audio-frequency decade resistance box for tests at lower

currents. The results are shown in table 3 and are estimated to be accurate to better than 0.01 percent.

A positive sign in this table indicates that with equal

Table 3. Results of comparisons of selected thermal converters with shunted model A thermal voltmeter

Thermal converter range

Test

current

Comparative ac-dc diflerencc

25 c/s 100 c/s 400 c/s

mn

50 ma

60

1 70

\ 100

240

200 400

600

400 600

I 800 amp

I 12

1 '°

1 20

I 24

/ 20

%

-0.003 % %

lUO -0. 043 -.062

200 .-- ---- +.002

-.065

-.010

-.044

-.056

-.068

-.041

-0.065 +.002

-.042

-.054 -.069

500

1,000

-.070

-.060 -.059

-.014

-.ois

amp

10

-.061

-.004 -.006

-.015

-.020

-.00«

-.017

-.024 -.010

-.010

20.. -

-.018

-.003

50

-.023

-.008

1 40 -.010 -.010

Thermal converter

ranges

1 to 2 ma

2 to 5 ina

5 lo II) Ilia 10 to 20 Ilia 20 to :!0 ma«

:«) to .SO ma M to KKI ma.

KKl to 200 ma

200 lo 5(KI ma

500 lo 1,000 ma...

1,000 ma to 1 amp

1 lo 2 amp

2 to 5 am))

5 to 10 amp

10 to 20 amp

20 to 50 amp

Tost

cur- rent

1. 2 2.4

6

12 24

.50 UO

HX) 240 000 amp

1.2

12

5

1020

Coinpiirativc ac-<lc diflerencc

2.S c/s

0.000 +.000 +.0O4 .000 + .001

100

c/s

+.004 +. 024 +0. 028 -.046 +.026

-.002

+.0)4

-.028

-. 123 -.008 -.010

-.014'

-.002

-.011 -.006

-.010

-.015

400

c/s

-o.oo:t

-.014

1,000

c/s

0. 000 +.(Xli .000 +. (KM -.001

.000 +. 031i -.044 -.002

-.005 -.002

+.00.3 -.006

-.008

-.018

10.000

c/s

-0.050 +.002

-.056

-.002

+.004

-.007 -.012

-.035

20.000

c/s %

+0.006 -.OOJ

.000 -.006

.000

.000 +.027

-.049

+. 039 +.00^

-.003

+.001 +.009

-.IXM

-.OOii

« This 30-ma thermal converter was the one used in the model A voltmeter.

169
response of the thermal converter on alternating and on direct current the response of the shunted model

A voltmeter was greater on alternating than on

direct current. The results of tests of the 1-, 2- and

5-amp thermal converters, duplicating those with

the electrodynamic ammeter, are not shown in the

table. The average difference between eight such

duplicate results obtained with this ammeter and

with the shunted model A voltmeter was 0.004 percent, and the largest difference was 0.013 percent.

A few additional check tests of these thermal converters with the shunted voltmeter were made at other frequencies but are not tabulated.

The results of all these tests were combined to assign reliable values of ac-dc difference, 5, to each

thermal converter in the series from 1 ma to 50 amp

and to the model A voltmeter. These values were

less than 0.01 percent for the model A voltmeter and

for all the thermal converters of ranges less than 100 ma. The values for the other converters (rounded

otf to the nearest 0.005%) are shown in table 4.

Table 4. Observed percentage ac-dc differences of thermal

converters {to the nearest 0.005 -percent)

(a) Differences independent of frequency

Converter

range

Percentage of rated current

40 50 00 80 100 120

ma

1 through 50.-.

100

(») (•) (•) -0. 030

(•) "-.'660

.000

(>) ()

-o.oeo

(•)

-.010

-.015 -.0.5

(») -0. 065

(•) -.070

-.016 -.020

-.025

200

500 -. 1000 amp

5

10

(*) -O.OiS -.040

.000

(•)

-0. 005 -.005

(•) -0. 055 -.060

-.005

20

50 -.010 "-.6i6

(b) Differences dependent upon frequency

Converter

range Testcurrent

Frequency (cycles per second)

25 00 100 200

amp

1

amp

fO.4 h.o

U.2

(0.8 ^2.0

12.

4

-0.0.0 -.055

-.065 -.020

-. 120 -. 155

0.000 0. 000 -.010

-.010

.000 -.010

-.015

o.oco

.000 —.005

.000

.000

.000

2

-.015

.000 -. 025 -.035

> Less than 0.01 percent throughout. The model B voltmeter, which was developed to give a \vider voltage range and greater convenience

of use than the model A, was compared with the

model A voltmeter at rated voltage on each range

up through the 300-v range and at 400 v on the higher

ranges, at selected frequencies from 30 to 20,000 c/s. The results are shown in table 5 and represent

directly the ac-dc difference, 5, of the model B voltmeter. They disclosed significant but not unexpected errors for the higher voltage ranges at the

higher frequencies.

Table 5. Hcsxdts of comparisons of model A and model B

thermal voltmeters

Rangr of model n

Voltage

ac-dc difference of model B voltmeter

25

c/s 1,000

c/s 5.000

c/s 15,000c./s 20,000c/s

Volts 3 through 75_.

150 300 600

V Rated .-. %

(•)

+6.003

%

(')

+o.'6ii

%

(-)

+0.614 +. 028

or y)

+6.044 +. 064

%

(') +0.013 +. 02i +. 05. +. 082

<lo do

400

400

« Less than 0.01 percent. 25 to 20,000 c/s each range,. As a result of these extensive series of tests, the

transfer performance of each of this first set of thermal converters and of each of the two voltmeters was uniquely determined. Small but significant ac- dc differences of both known and unsuspected causes were found. Corrections for these differences can

readily be applied when these thermal converters are used as transfer standards. However, an investigation of such differences is an especially important

objective of a development such as this. As a result

of such an investigation, a number of factors that

limit the transfer accuracy of an electrothermic in- strument were discovered and evaluated, with the

results given in the following sections.

5. D-C Response

In all measurements with a transfer standard the

average of the d-c response of the standard for the

two directions of current is taken as the reference upon

which the a-c measurements are based. For all three types of such standards, the reversed d-c aver- age gives a better basis than does the response for one

direction alone as effects that lead to small differ- ences in response for the two directions of direct cur- rent are reduced to the second order of smallness

when the average is taken.

Revei"sal differences may be expected in a contact

thermal converter because of the flow of heater cur- rent through the hot junction of the thermocouple.

This can cause a voltage drop in the thermocouple

circuit, which may lead to second-order errors be- cause of the nonlinearitics of the thermal and electric

circuits. This cause of reversal difference is minimized by the special constructions used in most

high-range thermal converters, and is eliminated by

the use of the small insulating bead between the

heater and the thermocouple in the low-range thermal converters. Surprisingly, however, early tests of

the first thermal converters purchased for the model

A voltmeter showed appreciable reversal differences

(up to 0.2%) even in tliese insulated-heater thermal

converters. It was soon realized that this was probably due to thermoelectric effects in the heater cir- cuit. Peltier and Thomson heating of the heater can

cause dissymmetry in the temperature rise of the

heater, and thus cause differences in the emf of the

thermocouple unless the hot junction is exactly at the

170
niiiipoiiit of tlio healer.* Tliis supposition was verified by an analysis of the data, which showed that

the ditrercnces were larger in those thermal converters

with heaters having larger thermoelectric effects. Further verification was secured by tests of special

thermal converters, constructed by one of the manufacturers " at the Bureau's request, with the hot

junction of each deliberatel}' and considerably off- center. The reversal differences were much larger,

and the signs and magnitudes were found to be

roughly in agreement with calculations based on

thermoelectric data for the alloys used.

Early intercomparisons of these few thermal

converters indicated that no significant ac-dc dif- ferences should be expected, even with rather large

reversal differences from this cause, provided that

the average for the two directions of current is taken as the d-c response. However, intercomparisons and tests of the much larger number of

thermal converters later purchased for current

measurements showed that a good proportion had

measurable ac-dc differences, all of the same sign

and independent of frequency. This suggested

a d-c error, for which these thermoelectric efforts seemed the most likely cause. This was investigated theoretically (see appendix 2) by considering the temperature rise of a homogeneous con- ductor heated by an electric current and cooled

solely by conduction to two relatively massive

terminals; i. c. by neglecting, for simplification,

radiation and convection losses and the heat abstracted by the attached thermocouple.

With only the normal joule, or resistance, heating,

and with the terminals of the conductor at equal

and fixed temperature, the temperature distri- bution along the conductor is parabolic, and the

temperature rise at the midpoint is, as Goodwin

showed,'" equal to ZL

8pk' (1)

where Vc is the voltage drop across the conductor,

and p and k are the electrical resistivity and thermal conductivity, respectivel3^ In addition, however, there is Peltier heating or cooling at the junctions of dissimilar metals (the two junctions between

the heater and its two supports) and Thomson

heating along each half of the wire. Unlike joule

heating, these are dependent upon the direction

of the current flow and can thus cause a dissymmetry of the temperature distribution along the

conductor, which revei-ses when the direct current

through the conductor is reversed. On alternating

current, even at the low frequencies in which we

are interested, the reversal of current occurs so rapidly that the thermal inertia of the wire prevents

any such dissymmetry, and the temperature dis-

' This dissymmetry has actually bcoii utilized to measure the Thomson effect. See (9].

• M. Roscnfleld, of the Field Electrical Instrument Co., who first showed that

the reversal differences were dependent upon the type of heater material. See (6). In addition to this classic paner, Goodwin has written an excellent

?eries of articles on "Thermal problems relating to measuring and control services", which aopeared in Weston Eng. Not^s, 1948-50 (Weston Electrical Instru- ment Corp., Newart, N. J.).

tribution is unchanged by these thermoelectric

effects. Calculations show (see appendix 2) that

with direct current the temperature rise at the

midpoint of the conductor, where the hot junction

of the thermocouple is attached, is unaffected by

the Peltier heating, for completely symmetrical

construction of a thermoelement. They show,

however, that it is changed by Thomson heating,

and becomes, approximately,"

(2)

where dt is the midpoint temperature rise in the

presence of Thomson heating, in ° C; da, the rise without Thomson heating; a, the Thomson voltage

coefficient in volts-degree Vc, the voltage across the conductor in volts. This was derived by considering a a constant,

whereas both experimental results and the electron

theory of metallic conduction indicate that a=BT,

where 5 is a constant and Tis the absolute temperature. However, a solution by a perturbation

m.ethod,'^ of the nonlinear equation that results when

a=BT, shows that the above results give a suffi- ciently close approximation for the temperature rise of about 200 deg C encountered in typical thermal

converters. Thus eq 2 shows that an ac-dc difference, or transfer error, should be expected in a thermal

converter, and enables the approximate calculation

of tliis error. Such calculations have been made for som.e comm.on metals and some of the alloys used as heaters in thermal converters, on the assumption of a 200 deg C rise at the m.idpoint, and a voltage drop

of 0 .2 V at rated currents." The results, converted to ac-dc difference for the same temperature rise, rather

than tem.perature difference for the same current, are shown in table 6. For the computation of these

results, the Thomson coefficient a was considered to be equal to i?(eo+2/3&d+273), where B is the second

derivative of the characteristic curve of emf versus temperature for a thermocouple composed of the

indicated metal and lead.''' is the ambient temperature in deg C. The results show approximately the

errors to be expected for these materials.

Table 6. Calculated ac-dc difference at rated current due to Thomson effects in thermal converter heaters

Metal B

ac-dc

difference

Percent Copper... ... +0.8X10-" <0 005 Platinum -3.2 -.013 Manganin , -0.08 <.0O5 Constantan (60 Cu-40 Ni) -7.3 -.064

90 NI-10 Or. _ -3.2 -.013

85 Pt-15 Ir -2.1 -.005

" At other points along the conductor the average temperature rise tor the two

directions of direct current is also changed by Thomson beating. " This method was suggested by Chester Snow. " From eq 1, a series of thermal converters having similar thermocouples and

the same rated output emf at rated current, and having heaters that olwy tbo Weidemann-Franz law will all have the same temperature rise and the sam«

voltage drop across the heater. The temperature rise is about 200 deg C, ai>d the

voltage drop about 0.2 v at rated current for most thermal converters.

I* This weighted average absolute temperature gives results in closer agreement

with those computed by the more exact method than does the unweighted average. M-l/2»i+273.

205001—52- 171
Tlie results may be conipared witli the observed

ac-(lc differences sbowni in table 4. From infoi ination

supplied by the mantifacturers, the 1-, 2-, 5-, 10-, 20-,

50-, and 200-ma tbermal converters have heaters of

either carbon, nickel-chromium, or i)latinumrhodium-ruthenium alloys. The observed ac-dc dif- ferences of these converters were less than 0.01 ])er- cent. The 100-, 500- and 1,000-ma thermal converters had heaters of constantan (Advance). 'J'he observed differences for these converters (at rated

current) were close to the calculated value of —O.OG

percent. The 1- and 2-am.p converters, of platinum.-

iridium, had observed differences (from table 4,b)

less than 0.005 percent, except at low frequencies,

also close to the calculated values. The 5-, 10-, 20-

and SO-am.p converters, of a special ])latinum-3-

percent-copper alloy, had difl'erences of —0.010 to —0.015 percent. Thus the agreement between tlie computed and observed values serves as excellent

verification of this source of error in transfer thermal

converters. Very fortunately, the error is significantly large only for materials having imusually

large therm.oelectric effects, such as constantan.

However, this does lead to the interesting conclusion

that for operation at a low temperature a thermal

converter should have large thermoelectric effects in the therm.ocouple, but that for the highest transfer

accurac}^ it should have very small thermoelectric

effects in the heater.

Additional experim.ental verification of this cause

of transfer error was secured by constructing'^ at the

Bureau two 5-anip therm.al converters tliffering only

in that one had a heater of manganin and the otlier of

constantan. These were 5-amp thermal converters

\\'ith a small Borax bead between the hot junction of a copper-constantan thermocouple ami the center of

the heater wire, to eliminate contact effects. Careful

ac-dc difference tests with the standard electrodynam.ic ammeter as the reference established agreement within 0.005 percent for the m.anganin element

and indicated an ac-dc diflerence of —0.06 percent

for the constantan element at an emf of about 5 mv. As a result of this work, a second set of thermal

converters in the 1- to 1,000-ma range was purchased

under specifications proliibiting heaters of constantan

or similar alloys. All of these converters w^ere found

to have ac-dc differences less than 0.01 percent at audio frequencies.

6. Low-frequency Response

It has long [10] been knowni that there is a lowfrequency limit to the correct rms response of a thermal converter, for reasons analogous to those causing a similar low-frequency limit in all other types of

transfer instruments. Below this limit the response

characteristic E=/(I) is not independent of frequency

(except for symmetrical square waves), and may be

expected to be in error by an amount that is inversely

proportional to the square of the frequency. As

table 4, b, sliows, such an error was observed for the

1- and 2-amp thermal converters.

" These thermal converters were constructed by E. S. Wilhams.

Wiien a constant direct <'urrei)t is iii)])Ue(l, the

iieater of a thermal converter reaches a constant

temperature. When an alternating current is ap-

])lied (after the transient state is over) tiie tem])erature of the heater varies cyclically about a mean

value, with an ami)litude that decreases with increasing frequency because of the thermal capacity of

the elem.ent. At frequencies high enough so that the

cyclic variations are negligible, the tem])erature rise of the heater is independent of frequency (])rovided

that at very high frequencies tlie skin effect and

electromagnetic radiation are negligible and that

standing waves do not alter the temperature distribution). However, at lower frequencies the mean

value, to which the d-c instrument or potentiometer

connected to the thermocouple responds, may not be

the same because of nonlinearities in the thermal and

electric circuits. These nonlinearities are caused

])rincipally by radiation losses, the dei)en(lence upon

temperature of the electrical resistivity and the thermal conductivity of the heater, and the curvature of

the em.f versus temperature characteristic of the

thermocouple. A theoretical evaluation of these ef- fects for the conditions in a thermal converter, in

which most of the cooling is by conduction to the

heater, is difficult, but does serve as a guitle in determining the approximate limiting frequencies for an

arbitrarily chosen error and for determining which

factors influence that error. Such an ap])roximate

analysis has been carried out by a perturbation

method (see appendix 3). The results show that the

average (time average) temperature rise at the mid-

])oint of a conductor heated by a sinusoidal electric

current and cooled solely by conduction to massive

terminals is approxim_ately

»ac = ^.c[l+|0.c2'] (3)

and that the percentage ac-dc diflerence of a thermal

converter having the hot junction of a thermocouple

fastetu'd to the midpoint of such a heater is approximatelv

h=-2bJIdac<r (4)

where

0„e=the midpoint temperature rise in degrees centigrade on alternating cur- rent of rms value / amperes

0rfc=the midpoint temperature rise in degrees centigrade for the same value

of direct current. h = a—l3—2NTll^, wiiere a is temperature

coeflicient of electrical resistivity.

l3 = the temperature coefficient of thermal

conductivitv of the conductor,

2>anda = the perimeter and area of the cross section of the conductor, respectively, f =thc einissivity.

/v = the Stefan-Boltzmann constant in watts-cm~--(deg IC)'*

h = thc thermal conductivity in wattscm^'-deg~'

172
7'o— tlu' ;iinl)ici)t toinpcrn t uic in degroos

Kelvin

q—d/ojl'^ and is (approximately) the ratio

of the crest value of the cyclic

fluctuation of temperature at the

midpoint to the averajjjo temperature

rise at the midpoint

aj = 27r/, where / is the frequency of the

applied current in cycles per second.

(/=the thermal difl'usivity of the heater

material in cnr'-sec"'.

Z = one-half the length of the conductor in centimeters.

TI=h-\- B/2A, where A and Ji are the constants in the emf-temperaturc characteristic for the thermocouple it- self, that is, in the expression,

E=Ad^{I}l2)d'

These equations indicate the factors upon which

the low-frequency error depends. They show that

the temperature rise can cojiectly be used as a measine of the jins alternating current, at fre- quencies high enough so that the second term in

the bracket of eq 3 is negligible for the accuracy

desired; in other words, at frequencies for which

the heater integrates reasonably well. This is true even if the thermal and electrical conductivities

of the heater are dependent upon temperature.

As a corollary, these equations show that a thermal

converter correctly indicates nns response, except

at low frequencies, even though its response characteristic E=f(I) is not quadratic.

The above equations are valid ojily at frequencies for which the second term in the brackets of

eq 3 is small compared to 1

.

Equation 4 may be used to calculate approximately the fiequencies for which ac-dc diiferences

may be expected to cause an error of an arbitrarily

chosen value. This has been done for some typical

materials for an error of 0.1 percent, a temperature

rise of 200° C, and a typical heater length of 0.4 cm. The results are shown in table 7.'®

Table 7. Frequencies for tuhich the computed ac-dc differ- ence is 0.1 percent

[TC-A signifies a thcnnocouple of Chromol P ami .Miimol, mid TC-B a thermocouple of eopi>er and coiistantan, forwhicli 1)/2A = -II.1X1U-' and +0.yxI0-3,

respectively]

"The effect of radiation losses has been neglected in preparin? this table. In general, this effect is significant only in the very low and very high range

thermal converters.

It is evi<leiit that the jnost significant factor iii the error term of eq 4 is the length of the heater

as tlui error is depen(Jent upon the fourth power

of this (|uantity.'' It is of interest to note that

the lengths, 2/, of the heater of the 1- and 2-amp

thermal converters are only 0.28 and 0.19 cm,

respectively, and that the length of each of the

higher-jange thermal converters is about 0.5 cm. The value of H can be determined from the departure from "square law" of the chaiacteristic equation E=J{r) of a thermal converter. For these

thermal converters it was considerably less than

the factor given for platinum in table 7. The

estimated frequencies for a 0.1-peicent enor, on

the assumption that the diffusivity of the platrinum-iridium alloy used in these thermal converters

is the same as that of platinum, were about one-lialf

the observed values.

7. High-Frequency Response

As these thermal converters are inherently suitable at frequencies up to 100 Mc or more, the upperfrequency limit is set by the circuits in which they

are used. The precautions indicated in section 4

arc sufficient to control capacitance and leakage

currents in tests of ammeters at frequencies consiilerably higher than the 20,000 c/s established

as the present upper limit by the demand for this

testing service. The actual upper-frequency limit

has not been established.

The magnitude of the effective impedance sets the

upper frequenc\^ limit for the model A and the model

B voltmeters. For an accuracy of 0.01 percent at 20,000 c/s, a time constant of less than 1 X 10"' and a

skin effect less than 0.01 percent are necessary for the impedance of each step of tlie model A and each

range of the model B instruments. Tlie time constant of each decade of the model A voltmeter was

computed from the inductance and direct-capacitance values furnished by the manufacturer of the

resistance box used. This was less than 5X10~* for

each decade. The skin effect was also negligible for

each decade. The resistance box is connected as a

three-terminal resistor, with the shield connected to the grounded side of the line. For this connection

the errors caused by the capacitance currents to the

shield are much less than those that would occur if the shield were connected to a resistor terminal.

They are significant only for the two l,000-ohm&-

per-step decades. The cfTects of the capacitance of

the resistance cards and switch studs of these decades

to the shield were computed from the manufacturer's capacitance figures of about 10 ^^f pcr decade

by considering the capacitance as distributed along a uniform transmission line, composed of resistance

elements and short-circuited at the end. From the

usual transmission line formulas, the magnitude of

I' For a given heater alloy, th" low-lreqiieiicy range r.ui Im' extended by using

longer heaters, with pr»|)o'tionati'ly gre.iter (Tos;;-seetional area to give the same

tomiM'ratnre i ise fnr the same current. If solid tioators are used, the improved

low-fro(iuency ix^rfornianec will be acoompanied by increased error due to skin

effect at high frequencies.

Heater material

Ap()ro.\imate temperature

cootficient Thermal

dillusivity

Frequency for i=0.1«

Hlcctrical

resistivity Thermal

conductivity TC-A

m

i23

32

5

Tc-n

c/s70

142

2

2

1

Cupix-r Platinum Maneanin.

Constantan

80 Ni-20 Cr. Carbon

+.'}.yxio-»

+3..'> < .1 < .1

. 1 —.ft

%rc . -0. 2X 10-1 5 +2. 7 4-2.3 '•+3.5 + .4

cm^-sec-'

I. 1

. 2.'> .075 .WA

.038

^ 2

• For a heater 0.4 cm long with a temiKTature rise of 200° C. Estimated from Lorenz' I.,aw.

173
the current at the sliort-circuitcd end was computed

as approximately

1/1-^ [1 -0.0056 (aj(7/?n, (5) h

where

l'=the applied voltage

/ = tlie current at the short-circuited end

/?=the resistance in the circuit

C=the total capacitance to shield of the

windings in use

At 20,000 c/s for a capacitance of 30 ^^f this is changed by less than 0.005 percent at 20,000 ohms,

the maximum setting. The effect of the capacitance

(about 10 ^l^tVj of each rather large switch blade to the shield was separately computed, and found to be

less than 0.005 percent at 20,000 c/s. Thus it was

concluded that the errors of this voltmeter should

be a maximum of 0.01 percent at the highest settings

and frequencies used, and less at lower settings and

frequencies. The performance at the top settings was verified by the tests outlined in section 4. The impedance of the model B voltmeter was similarly computed, and only the capacitance to the shield was found to be significant. As a rough but adequate approximation this may be considered the

same for each step, which leads to the approximation of considering the capacitance as distributed

along a nonuniform transmission line composed of

resistance elements and short-circuited at the end.

The effective impedance was computed by a perturbation method,'* on the rough assumption that the

resistance per unit length was proportional to the

square of the electrical distance from the output end.

The magnitude of the current at the short-circuited

end was found to be approximately

|7|=Z[i_o.0022(a;(7i?)2]. (6) H

From the measured capacitance of 50 mxi and the

total resistance of 100,000 ohms for the 750-v range,

the change in current was computed as —0.09 percent at 20,000 c/s, which is in very good agreement

with the observed results of table 5. Moderately

good agreement for the other ranges was also obtained, but the change in current was not observed

to increase in proportion to the square of the fre- quency as indicated by eq 6. However, the agreement is good enough to show that the ac-dc difference

of this instrument is due to capacitance currents to the shield. These currents cannot easily be reduced in a shielded multirange instrument of reasonable size. (A shield is desirable to control the currents.) They

could be compensated by connecting appropriate

capacitors across sections of the resistors, but this seems less desirable than the procedure of applying

small corrections and has not been done in the model

B voltmeter. Much of the capacitance is that

between the binding posts of each range (see fig. 2)

"See appendix *.

and between the high-side binding post and the top

shield. This shield is brought as close as 2.5 cm to the row of high-side posts.

8. Other Effects

The intercomparisons of insulated-heater thermal

converters disclosed an unsuspected ac-dc dift'erence

that was dependent upon the voltage between the

heater and ground and upon the heater current of

each of these elements. This was traced to a very

small leakage current flowing through the ceramic

bead between the heater and thermocouple and then

through the stray capacitance of the Lindeck potentiometer to ground. This current caused additional

heating of the bead and thus an error as on direct

current the insulation resistance of the Lindeck was much higher than the capacitive reactance, even at 20 c/s. In addition, the ceramic bead of each of

these thermal converters exhibited marked dielectric absorption, so that its a-c resistance was niuch

less than its d-c resistance. For a tj'pical thermal

converter at rated current a leakage current of 0.7 /na resulted from a difference of 50 v between the heater

and ground, resulting in a power dissipation of 35 n\v

in the bead. This was sufficient to cause a 0.2-percent error, which was proportional to the square of

the voltage difference and was very markedly dependent upon the heater current.'^ The reactance

of a capacitance as small as 100 ;u/xf is less than the

computed bead resistance, even at a frequency as low as 50 c/s, so that in almost any application this source of error makes it necessary to maintain the

heater near ground potential.

9, Other Applications

This equipment was designed for the testing of

other instruments. Portability and the ease of making other a-c measurements were not considered to be important. However, the thermal converters and

voltmeter elements are readily portable and have

been used, with a small portable millivolt potentiometer as the indicator, for measurements at other

locations. For measurements of current or voltage

to 0.01 percent, it is necessary to use a transfer

method, calibrating the thermal converter or voltmeter on direct current at the time of use. For a more moderate accuracy of about 0.1 percent, it should be feasible to use a low-range thermal converter, shunting it for current measurements and

adding a series resistor for voltage measurements.

Only the thermal converter itself need be transferred

to direct current if the resistances of the shunts and

the series resistor are properly adjusted. Thus d-c sources of extended range, volt boxes, etc., should be

unnecessary. The whole apparatus could be built

in a convenient case, containing a small potentiometer for the measurements of the current through,

or the voltage across the thermal converter heater.

i» This was due to the marked decrease in the resistance of the ceramic bead

with an increase in temperature. With a fixed voltage iicross the bead, the error was in some cases proportional to the tenth power of the heater current.

174
A suitable low-range thermal converter should also

make feasible an a-c potentiometer with an inherent

accuracy of 0.01 percent. The basic circuit for a

polar form of such a potentiometer (to measure the

magnitude but not the components or phase angle of an alternating voltage) is shown in figure 6. With

the switch thrown to direct-current the potentiometer is standardized in the usual manner, and the

deflection of the indicator of the thermal converter is observed. The switch is then thrown to alternating

current, which is adjusted for the same deflection

of this indicator. The circuit is surely not new, but

a proved thermal converter makes possible a con- venient transfer with accuracies comparable to those

of a d-c potentiometer. The other basic limitation

remains that any a-c potentiometer measures only

the fundamental component of the unknown voltage.

However for an alternating quantity having a harmonic content as large as 2 percent, the rms value

differs from the fundamental by only 0.02 percent.

Audio-frequency oscillators and amplifiers with amplitude distortion much less than this are now available. Such an oscillator could be used to supply the

potentiometer directly and to supply the test circuit

through an amplifier preceded by a suitable phaseshift network.

10. Summary

The electro thermic transfer standards used at the

Bureau for the precise measurement of voltage and

current at audio frequencies, and designed primarily

for the standardization of electric instruments, have

been described. The extensive tests to establish the

transfer performance of the standards have been outlined, and the results have been given. The causes

of the observed ac-dc differences have been outlined

and the errors evaluated. The equations governing

the temperature rise of a conductor heated by an

electric current and cooled by conduction to relatively

massive terminals have been solved to establish the

steady-state midpoint temperature rise on direct

current with Peltier and Thomson heating, as well as the ordinary resistance heating, in order to obtain

the d-c error of a thermal converter. The equations

have been solved to establish the average midpoint

temperature rise on alternating current, in order to obtain the low-frequency error of a thermal converter

in which the electrical and thermal conductivities of

the heater are temperature dependent, in which some

loss of heat occurs by radiation, and in which the

characteristic of the thermocouple is nonlinear.

Possible applications and modifications of the equipment for measurements under less stringent conditions have been suggested, and the application to an a-c potentiometer of excellent inherent accuracy

has been diagrammed.

The results of this work establish the excellence of

thermal converters as transfer standards of highest

accuracy, comparable over wide ranges with that

obtainable with other types of standards at much

narrower ranges. The requirements for such standards are feasible and the manner of use to insure such

accuracy not unreasonable. The c^st of the required

number of standards and the equipment is not prohibitive. Thus electro thermic instruments may be

ranked on a par with electrodynamic and electrostatic instruments for a-c measurements of the highest accuracy.

As a result of this work, thermal converters of

negligible or small known transfer error are now

available as transfer standards with whicli tlie transfer performance of other thermal converters may be

evaluated at the NBS to 0.01 percent at audio

frequencies.

The author acknowledges the help of Murray Blitz,

who made most of the intercomparisons described in

section 4 of this report and checked all the tables and

equations, and of Earl Williams, who constructed ail the equipment and assisted with the measurements.

11. References

[1] F. K. Harris, A suppressed-zero electrodynamic volt- meter, BS J. Research 3, 445 (1929) RP105.

[2] J. H. Park and A. B. Lewis, Standard electrodynamic wattmeter and ac-dc transfer instrument, J. Research

NBS 25, 545 (1940) RP1344.

[3] F. B. Silsbee, Composite-coil electrodynamic instruments,

BS J. Research 8, 217 (1932) RP411.

[4] C. C. Patterson, E. H. Rayner, and A. Kinnes, The use

of the electrostatic method for the measurement of power, J. Inst. Elec. Engrs. (London) 51, 294 (1913).

[5] R. S. J. Spilsbury and A. Felton, The electrostatic volt- meter as a dc-ac transfer instrument, J. Inst. Elec. Engrs. (London) 89-11, 129 (1942).

[6] W. N. Goodwin, Jr., The compensated thermocouple

ammeter, Trans. Am. Inst. Elec. Engrs. 55, 23-33

(1936).

[7] F. L. Hermach, A precision electrothermic voltmeter for measurements between 20 and 20,000 cycles, Trans. Am. Inst. Elec. Engrs. 67, 1224 (1948).

[8] H. B. Brooks and A. W. Spinks, A multi-range potentiometer and its application to the measurement of small temperature differences, BS J. Research 9, 781 (1932)

RP506.

[9] G. Borelius, Bestimmung des Thomsoneffektes aus dem

thermischen Gleichgewicht in einem str5mdurchflo3- senen Drahte, Ann. Physik 63, 845 (1920).

[10] M. G. Lloyd, Elec. Rev. & Western Elec. 58, 65 (Jan. 14, 1911).

Appendix 1. Development of Working

Equations

By definition, the percentage ac-dc difference of an instru- ment is the percentage difference in the quantity required

to give the same response on alternating and direct current.

It is 8=100 (7)

where Q is the quantity (current, voltage, or power) that

the instrument measures.

In a "straight a-c test" the direct current or voltage is adjusted to give the same response of the transfer standard

observed on alternating current. The direct current or

voltage is measured with the potentiometer and its accessories. Therefore, from the above definition,

= (l +4). (8)

175
where Ode is the quantity measured with the potentiometer,

5, is the ac-dc difference of the transfer standard, and Q„r

is the a-c vahie required for the observed deilcction of the

test instrument.

Thermol

Converter Figure 6. Elementary form of Polar a-c polenliometer.

When a thermal converter is used for transfer tests tlic small percentage difference. Si, between the values recjuired for the same indication of the test instrument on alternating

and direct current is determined from the resulting deflection of the Lindeck galvanometer. Therefore, for each transfer thermal converter, the small percentage difference in output

emf resulting from a small percentage difference in heater current must be evaluated. This relationship may be

expressed as -^= ny. (9)

where E is the output voltage, / the lieater current, and n

a number that is generallj' close to 2, but may range from

from 1.5 to 2.2. The characteristic, //, may be evaluated on

direct current by measuring with a potentiometer the small percentage differences in E res'.ilting from small percentage

differences in heater current successively set with a po- tentiometer at selected current levels, then computing n by the above expression and plotting n versus E. The

characteristic need not be determined with an accuracy

better than 1 percent and is relatively permanent.

If the Lindeck potentiometer shown in figure 3 is used to measure E and AE in millivolts in a transfer test, then at balance E= Vp,^" the voltage drop across the four-terminal

resistor. If the galvanometer deflection D, is directly pro- portional to the current through the galvanometer 2' and if the potentiometer voltage is unchanged during a determination,

where AD is the difference in galvanometer deflection corre- sponding to a difference AE in the applied emf, Rp is the

resistance of the galvanometer circuit, and iS is the current

sensitivity of the galvanometer in centimeters per milli- ampere on the scale. In the Lindeck potentiometer used in this setup, the galvanometer sensitivity is adjusted so that numerically

100jgp

S

These relationships may be combined with eq 9 to give

(10)

An accuracy of 1 percent is suflficieut for this incasuroiiit'nt. " The use of a straight instead of cylindrical scale introduces an error that can be shown to be less than 1 percent if the palvanonicter-to-scale distance is at least three limes the scale length.

Therefore, for a transfer test of an ammeter with a thermal

converter that lias no transfix error, tin; required ac-dc

difference of the test instrument is by vq 7 and 10

where the alternating and direct currents ari; adjusted for the .same respon.sc of the lest instrument, and the resulting

galvanometer deflections are D,,^ and J),/,, respectively.

If the transfer thermal converter has a ])ercentage ac-dc

difference, 5,, at the current and freciueiicy at which the lest is made, then the ac-dc diffiTenci; of the instrument under test

is, to the desir(>d order of ap])roximat ion.

Similarly, for a voltmeter test,

in which 5, is the percentage ac-dc difference of the range (or setting) of the standard voltmeter used. In this deflection method of using the Lindeck potentiometer for rapid transfer tests, the resistance of the circuit seen by the galvanometer must be constant to better than 1 percent. The resistance of the thermocouple of each thermal

converter has been adjusted to the same value .so that the

simplified formulas may always be used. The temperature

coefficient of the resistance of the Lindeck circuit has been

computed as less than 0.1 percent per deg C, and computations

show that the resistance seen by the galvanometer is changed

by less than 0.2 jjercent by the adjustable battery circuit, for

all ranges of the i)otentiometer cxcejit the tt)pmost.

Appendix 2. Effect of Peltier and Thomson

Heating

The problem is to find the steady-state temperature rise of a uniform conductor of length 21 and uniform cross-sectional

area, a, carrying a constant current, /, and cooled solely by

conduction to its terminals. The joule heating of a length dx occurs at the time rate I-pdx/a watts, where p is the electric resistivity in ohm-cm. The differeiice in the heat conducted

per second across the two ends of this differential element is ak{d''0/dx-)dx watts where ^ is the thermal conductivit}- in watts-cm~'-degrees"', and 6 is the temperature of the element in deg C. Equating these and dividing by dx gives the differential equation governing the temperature of the conductor

ak r-;-r- =0,

dx^ a

(14)

which is of the form

e"=-h, (1-la) where b= Pp/a^lc.

If the terminals remain at equal temperatures ^o, and if the

origin is at the midpoint of the conductor, the boundary conditions are 0 = do at x= ±1. The solution, by direct integration and application of these conditions, is

6- and at the midjwint

-^0—2—

«

(15)

If tlie ternunals are of different metal than the conductor,

Peltier heating and cooling (proportional to the first power of the current) may occur at the junctions of the conductor with

the terminals. These will be equal and opposite in sign at the two terminals, and, if we can a.ssume complete tliermal symmetry, their effect will be to raise the tenipcrtaure of

176
one terminal by a small amount and to lower the other by

an eqiial anionnt. Tlie different iai etnialion then ha?! as its boundary conditions, 0=00+ AS at 2=+/ and d~di,- AO at x=—l. Th(! solution is

h,,., , AOs 00=2 (HI)

and is therefore niichanf^ed at the niidi)oint Of course,

coin()lete therma! symmetry is never attained, so that I'eltier heals may cause a chanf^(? in the midpoint temfierat ure. Ho\ve\ i'r, for massive termitials the teni])eral ure changes,

A8, are \ery small, so that Peltier heating need not be considered significant.

TJiomson heat iiig occurs in homogeneous conductors having

a temperature gradient in the direction of the current flow. For the one-dimensional case here considered, the Thomson

heat in an element of'lengtJi dx having a temperature differ- ence di/ is a /do watts where a is the Thomson voltage coefTicient. pjxperiment aii(t theory indicate that a is proportional to the

ab.solute temperature, but, as tlie Thomson heating is small compared to the joul(> heating, we might, ay a first approximation, consider a constant. Equation 14 then becomes

,d^,Pp, .do „ dx^ a dx which is of the form

r' + cO' + b^O

(17)

(17a)

with tlie boundary conditions 0 = 8o at x= ±1, where c=al/ak.

The steady-state solution of this eq is

c \tanh q sinh q/ c

where 9 = c/<<l. At the midpoint, where x = 0

q \ smh q /

Expanding the hyperbolic functions gives, by neglecting

higher-order terms,

^.-^o = 0, =^X'-©='^(^-©- ^'^^

By substituting the values for b and c and noting that

2Ipl/a = V, the voitage drop across the conductor, eq 1 and 2

of the text can be obtained. They show that Thomson heating causes a small (second order of smallriess) decrease in the temperature rise at the midpoint of the conductor, and can

thus lead to a transfer error in a thermal converter. However,

the approximation of considering a a constant is a drastic one, and a more accurate approach will be to consider a=BT,

where B is a constant,-' and T is the absolute temperature.

Equation 17 then becomes

ax' a dx

which is of the form

T" -{-wTT -\-b = Q, (1«J)

with the boundarv conditions T=To at x=±l, where

w=BI/ak

This is a nonlinear equation and very diflicult to solve. Chester Snow showed that it could be reduced to a Ricatti and

then to a Bessel equation, but suggested an approximation or perturbation method to give the desired results more simply

and directly. The answer is not significantly different from

J

Since PcltuT heating depends upon Itic junction temperature, a secondorder cliRnge in midpoint temperature may occur. B^tPEidV, wlierc /-,' is the enif of ;i I li(Tmoeuii|>le composed of ttie mel;d in 'lucstion and a motal sucli as lead, which has ver>' small Thomson heatinjz.

that of eq 18. The metliod is powerful, how(!ver, and was

used later in r'valualing other perturbing effects, so tfiat it will b(M)Ullined here. 'I'lu' Thomson healing, which is much

smaller 1 han 1 he Joule heat ing, can be consiflcred as pert urbing

tlur temi)eralure distribution of the conduclor. Its c;lfect can

b(! a()proximated by .solving th(; differential e()uation without

the perturbing term, then substituting tin; r('sult, and its deri vat i V(,'s as necessary, in the [xtI urbing tcrni only, and

.s(jlving the resulting e(|uation.

Without the Thomson heat t(!rm, ef| 1!) becomes

T"=^-b

with the boundary condition given, the solution of which is

where the subscript 1 inflicates the first approximation.

Placing this value of 'A and in the perturbing term of 19, reduces this to a linear equation,

r:= -b-~^^^'-Vwbx{T a+r,)>

where Td=bL-/2, the solution of which is

At x = 0 this reduces to Ty~Ta as only first-order terms are

given, so that the process must be repeated to find the secondorder term indicated by eq 18. The algebra ra[)idly becomes

very lengthy and tedious, but the procedure is straightforward,

and the resulting temj)erature rise at the midpt)int is found

to be aj^proximately

7^-ro=Trf[l-3.4X 10-^23- 10.5X10-22oZd-8.3X10-22iil,

(20)

where Zo=wTol, and zj=wTdl

Appendix 3. Low-Frequency Error of a

Thermoelement

The problem is to find the steady-state average (over an

integral number of cycles) temperature rise at the center of a uniform conductor of length, 21, and cross-sectional area, a, carrying a sinusoidal current and cooled chiefly by conduction

to massive terminals that are at fixed and equal temperatures.

Assume that the current remains sinusoidal and that only

Joule heating of the conductor occurs. If all the cooling is by

conduction to the terminals, the differential equation governing the teinpt^raturc of the conductor is, under these assutnptions,

ak = ox- a

do

-ains -r-^ at

(21)

with the boundary condition 8{±l,t) =0, where

0 = temperature in deg C

2' = the square of the instantaneous value of the current =

/-'(I -cos 2U)

/ = rms value of the current

ui = 27r/, aiul / is the frequency of the current in cycles per second

t — t ime in seconds

«( = density of the conductor in grams-cm~^

s = thermal cai)Hcitv of the conductor in joulcs-grams~'

-deg C-'.

177
The procedure will be to consider p and k constant and to solve for the periodic steady-state temperature rise, then to approximate p and fc as p = po (1 +a9) and k = ko (1+06), to in- troduce radiation loss, and to solve the resulting equation by

the perturbation method just outlined. For the cases considered aO and 06 are considerably less than 1, and radiation

loss is much less than the Joule heating of the conductor.

With a and ^ = 0, eq 21 is the form — 6(1 -cos 2wt)

where g=ms/ko, and b= Ppo/a^koLei e=u{x,t) + x('(x). Then

Let dV/dx2=-6. Then ^= - (6z')/2 + Cix+C2. Applying boundary conditions gives

e{ + l,l)=ni + l,t)+i(l) =

e{-i,t) = u(-l,i) + ^{,(-l)-

We mav take ^(l) = ^(-l) = 0. Then ^ = 6/2(P-x2) and eq 22 becomes

= 0.

-g -^ = 0 cos 2u)/, dx^' dt with m(±/,0=0.

From complex number theory if n is harmonic in time, it can be represented by the real part of Ue>^"', where U is a complex function of the real variable x, and j=-\J—l. Similarly, b cos 2wt can be represented by the real part of be'''"', where 6 is a real number. Substituitng these quantities and

the appropriate derivatives in eq 23 and dividing by e'^"' results in ^-j2.,gU = b,

with U=0 when x=±l. The solution of this is

(24)

H2 L cosh filj (25)

where n = -\/j2it>g. At the midpoint

^ C^ cosh

b

2tog'

as at the frequencies we are interested in cosh /iZ> > 1 . Therefore, u = real part of Ue>^<'*= real part of —- (cos 2a)t+y sin 2u>l) \ « — sin 2<A}1. Zujg I Zwg

So that, to this approximation, at the midpoint

«««=^(l-^sin 2a,<^=ed(l-g sin 2a,0, (26)

where di=bP/2 and 9= l/wgP. Note that q is the crest value

of the cyclic temperature fluctuation divided by the average temperature rise at midpoint.

In general, U= Ua+jlh, where (J, and IJb are functions

of X, so that u= L'a cos 2u>t— Ut, sin 2ait and

ffi =2 (^'-^')+f^. cos 2wt-Ui sin 2o,t. (27)

Thus the average value over an integral number of cycles of the temperature rise is the same as that found in appendix 2

for the same numerical value of direct current, so that under

these conditions the temperature rise of the condtictor can be used as a measure of the current without ac-dc error. We shall now consider that p=po(l+aO) and k = ko(l + 08) and that some small fraction of the total heat is lost by

radiation. If p is tlie perimeter of the heater in centimeters,

then the heat radiated per second from the surface of a

differential length, dj, of the heater is

dn^=p!:K{T*-Ti)dx.

where f is the emissivity of the heater material, a numerical

constant, and K is the Stefan-Boltzmann constant in watts- cm~2(deg K)~*. T is the absolute temperature of the ele- ment; To is the absolute temperature of the surroundings.

Introducing these terms in eq 22 gives, as PO<Cl,

||-9(l-^<>)|J=-6(l-cos 2cot){l + r,e)+^{T*-Ti)

(28)

where 17 = 0-

(1)

(2)

(3)

The perturbing terms are + Pg9

-6(1

ak

dt — cos 2wt)r)9

As these perturbing terms are relatively small, the solution

will differ by only a small amount from that given by eq 27. Hence a sufficient expression for these small terms will be given by substituting the known values of the dependent

variables y from eq 27, and its derivatives as necessary,

in these perturbing terms. When this is done, it is found

that the first of these terms reduces to functions of x, each

multiplied by sin 7iwl or cos nwt, where n = 2, 4. The second

reduces to similar harmonic terms plus the terms — 6[»;P—

(ijC/„/2)], where P=(P-xi)b/2. The third term becomes

unnecessarily complex. It can be simplified by making the

further approximation that for this term

and

T-To = Or=Pil-q sin 2w<),

T*-Ti={To + 9ry-Ti^4erTl + 6ejTl, 9r<To

and also that k = k(,. With these approximations, this term

reduces to harmonic terms plus the terms

4NTIP + 6NTIP'(^ 1 + 2 )

where N=p^K/ako.

Thus eq 28 becomes

ffg=-6(l+,P-^") + 2A^r5P[2To + 3Pl + (f)] +

(29)

71 = 4 23(^1 COS nw< + (j„ sin nwl),

n = 2 where Fn and 0^ are known functions of x alone. We again let 9{x, t) —u{x, t)-\-^{x) and set d^^/di^ equal to the terms

independent of { in eq 29, so that

^ —^^(f„ cos ruji + G, sin n&><) (30)

n=2

178
For each frequency of tlie harmonic driving terms in tlie right-hand side of this equation, the solution of the equation

that vanishes at the end points will be of the form (7, cos nw/— C/fc sin nojl, where Ua and f/j, are functions of x. The

average value, over an integral numlxir of cycles of the lowest frequency, will therefore vanish. Since this is a

linear equation, the solution is the sum of such solutions and will therefore also vanish. Thus the average value

of 9, which we are interested in, is equal to f (x). Evaluation <p (x) in the same manner as before gives for the value of ^ at the midpoint where x = 0, (+!")]

where

(31)

U„2{x)=^dx,^''Ua{x^)dXx

and on the assumptions that Uai(.—l) = Ua^Q) and (7aj(0) = 0. The required expression for can be evaluated from eq 25

and 27 by noting that

iix = Vj2wg x = yx +jyx and /iZ = +jyl,

where 7 = V^, so that cosh /xx=cosh yx cos yx+j sinh yx sin

rr, and cosh pi = cosh yl cosyl+j sinh yl sin yl. At the fre- quencies we are interested in so that

cosh >iJ»-2-(cos yl+j sin yl).

Since u=real part of

[2(og L cosh

it will be found that

cos 2U — f/ 4 sin 2w/,

Ua=- i [sinh yx sin yx cos 7Z — cosh yx cos yx sin 7/].

Performing the desired integration results in

b

From which

(cos yl cosh yx cos 71 -[-sin 7Z sinh yx sin 71).

C/o2(0 = f/-2(-0«-rViand Ua2{0)«U.2(l).

Therefore, eq 31 becomes

1^(0) =#<..=e<, [

1 -f M-f ATT?/') <?d9']=

ed[l-f M+ |0d3']. (32)

where M is equal to the remaining terms in eq 31 and A«7j —

2NTiP.

By the same procedure, eq 28 can be solved for the d-c case by considering dS/d<=0 and t =/, a constant of the same

numerical value as the rms value of the alternating current. The temperature rise at the midpoint will be found to be

0d.=ea[l + M]. (33)

Thus the temperature rise on direct current is the same as on

alternating current when the frequency is high enough so that

the third term in the brackets of eq 32 is sufficiently small compared to unity. Therefore the temperature rise at the midpoint on alternating current is

if M is small compared to unity.

If the hot junction of the thermocouple is fastened to the midpoint of the heater, the emf of the couple may be repre- sented by

where 0„ is the temperature of the hot junction in deg C, and

A and B are constants to be evaluated at the temperature of the cold junction. On direct current 0m = O<ic and

On alternating current of the same rms value, 0^ may be

considered to be closely enough for these purposes

<'« = 0d. [1+1 Od<</»] sin 2a,/).

The average value of emf over an integral number of cycles of alternating current is then (to the desired accuracy, and if {B/2A) die is small compared to unity)

From these expressions

Ea EA H

E4 2 where H=^h +B/2A.

For table 7, it is desired to compute the frequency at which

the ac-dc difference of a thermal converter due to this effect

is equal to 0.1 percent. From eq 7 and 9 and for n =2 the

difference in current for the same emf is one-half the difference

in emf for the same current, and is opposite in sign. Therefore the percentage ac-dc difference, 3, is approximately

&=~25Heacq^ -25Hei^

{2^f)H*

'

where in addition to the symbols already defined, d=l/g^

kt/ms and is the thermal diflfusivity in appropriate units. Appendix 4. Effect of Shield Capacitance

on the Model B Voltmeter

We may approximate the effects of capacitance to the

shield of this voltmeter by considering the voltmeter as consisting of a transmission line of rx=rox^ ohms per unit

length, having a constant capacitance co farads per unit

length to the shield, witha sinusoidal voltage of rms value V

volts applied to the high-resistance end. The other end may

be considered short-circuited, since the thermal converter, which is connected between the low-resistance end and the ground post, is of low resistance. The shield is also connected

to this ground post. We wish to find the magnitude of current, /o, at the short-circuited end. For an element of length dx &t & distance x from this end, we have

^=Irx = Irox^ and ^=jo}CoV

,

ax ax

(35)

where dT is the rms voltage drop across the element, and

dl is the rms current from the element to the shield. Differ- entiating the first of these and substituting the values of I and difdx in the resulting expression gives

d^V

dx^'

2 dV

X dx

\ixW =0, (36)

subject to the boundary conditions V = 0 at i = 0 and V =V,

&t x=l and where X'=ja,Coro.

When X=0 this becomes, with the same boundary

conditions,

d2V;_2 dV_Q

dx^ x dx '

179
the solution of which is

Wo now {)ut this vahic of V in tho jierttirhinn torni contaiiiiiig

X- in eq 36, to get

an'

dx2 dVX dx

Tlic ^olutio^ of this, with the same boundary corniil ions, is ^ P \ 28 J 2HP

As \-=jwCi,r(,, we will need to repeat this j)rocoss to securi' terms in X*, for we are interested in the magnitudes of V anri /. Putting the above value of V2 in the last term of e() 30 and

solving as before gives

+

X^F^t"

2464P

(37)

From ec| 35

I =—Tox^ dx (38)

J'erforming the indicated diflferentiation of ec| 37 and substituting in 38, and noting that tho total resistance is R = rjyS leads to

Rl 28 + 8.8X10-<X2/^^-

12 \ 28/ +

()72J

Solving for Iq at ^ = 0, and noting that the total capacitance

is C'= 0,/ gives

/„= ^"1^ 1 -jo. l07wCR - 7.9 X 10-3(coC/(' )']•

Since the terms in the bracket are small compared witli unity

|/ol =^"[^1-0.0022 (coC ye )2 J- Washington, August 16, 19ol.

180
Section 12

Thermal Current Converters for Accurate AC Current

Measurement

EARL S. WILLIAMS

Abstract—A new 14-range set of thermal current converters

consisting of shunted thermoelements has been constructed to measure ac-dc difference and ac current from 10 mA to 20 A at 20 Hz to 50 kHz. The ac-dc difference corrections for all ranges can

be determined relative to two ranges by a 7-step intercomparison

of certain adjacent ranges.

Introduction

THERMAL CURRENT converters (TCC's) consisting of ac shunts in parallel with thermoelements

(TE's) are widely used as ac-dc transfer instruments to determine ac-dc difference of other instruments and to measure ac currents by comparison with known dc currents. Shunt assemblies are sometimes incorporated into

multirange instruments, but usually are individual sin- gle-range units. They are attached to a thermal voltage

converter (TVC) which contains a low-voltage low-current

TE. Commercial multirange TVC's usually incorporate an

adjustable source of EMF and a null detector for monitoring the output of the TE.

Specific procedures and the necessary equipment have

not been provided by shunt manufacturers to enable the

user to make intercomparison tests between members of a set of shunts. However such tests would provide several

advantages. It would be sufficient for the manufacturer or

a standards laboratory to calibrate only two or three

ranges. These ranges would serve as reference points for calibrating the other ranges by intercomparison. Periodic

intercomparisons could then be made by the user to detect

any significant changes that might occur in any shunt in the set, and redeterminations of ac-dc difference relative

to other standards would not be required as often. The new NBS set of TCC's, consisting of six shunts and

two TVC's (Fig. 1) is designed so that it can be intercompared in a minimum number of steps. The TVC's have

nominal inputs of 0.3 and 0.6 V and may be attached to any

one of the shunts by a coaxial connector. This permits each

shunt to be used for two ranges, as shown in Table I where

current ratings, shunt resistances, and TVC currents are

listed. For the purpose of calibration, intercomparisons are made between each of the six shunts or between alternate

ranges. The TVC's are used without shunts for the two

lowest ranges (10 and 20 mA), and they are compared with

the shunted ranges in two additional steps.

Manuscript received June 28, 1976; revised August 12, 1976. The author is with the Electricity Division, National Bureau of Standards, Washington, DC 202.34.

Thermoelements as TCC's and TVC's

TE's without shunts have been used for some years at the NBS as primary current converters. They can be very

accurate ac to dc transfer devices. The short straight heater

and connecting wires have minimal reactance, and an in- sulating bead between the heater and thermocouple pre- vents any appreciable interaction between the current being measured and the readout instruments. The thermocouple output is rather low (7 to 12 mV), but it can be

monitored with very good precision with a modern galvanometer or microvoltmeter.

However shunted thermoelements are probably preferable as TCC's in most other calibration facilities mainly

because the low current TE's used are relatively inexpensive and easy to replace if burned out—an accident that can happen quite easily. Ampere-range TE's are more

expensive, and those with insulating beads are not ordinarily available. However, a few sets have been made by

one manufacturer on special arrangement.

TE's for 10 and 20 A usually have C-shaped heaters

(tubular except for opening on one side) which can cause an unusual effect. The ac-dc difference is very much dependent on the location of other nearby conductors, particularly the return conductor which is usually placed along

side of, and more or less parallel to, the TE heater. Apparently the ac current is distributed for minimum reac- tance and may therefore be concentrated close to, or away

from, the thermocouple which is attached to the top of the

heater. This nonuniform current distribution can cause an ac-dc difference change of 0.03 percent or more. Rigid

copper return conductors have therefore been added to these TE's and the input terminals are both at the same

end of the device. The conductors are also positioned for small ac-dc differences.

Similar TE's with straight wire heaters (0.5 to 5 A) and

insulation have ac-dc differences of 10 ppm or less at 50

kHz.

TE's for currents up to 1000 mA are available with in- sulation and vacuum bulb enclosures. They are adversely

affected by Peltier and Thomson heating, which have been

discussed elsewhere [1], [2]. However, skin effect is the

major concern in the practical use of these TE's. The

copper-coated nickel alloy widely used for passing current conductors through glass seals is sufficiently magnetic to produce a small increase in the ac impedance in the heater

support stems causing additional heating with ac current but not with dc current. The effect is dependent on both

181
520 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

TABLE I Thermal Current Converter Set

Fourteen ranges are fonned with six shunts and two TVCs (FG and FH)

.

Current Shunt TVC Range Resistance Current

(amperes) (ohms) (mA)

0.01 10 (FG)

0.02 20 (FH)

0.03

15 10 0.06 20

0.1

0.2 3.33

1020

0.3 10 0.6

1.0

20

1

2 0.3

10

20

3

6 0.1

10

20

1020 0.03

10

20

frequency and current thus complicating stepup intercomparison tests. A few special TE's have been made with nonmagnetic

Ni-Cr quaternary alloy for both the heater and supporting

stem inside the glass bulb. This not only minimizes the skin

effect in the conductors near the heater, but also eliminates

Peltier heating at the heater ends since the supporting

conductors and the heater are of the same material.

The heating due to skin effect is usually negligible at currents less than 200 mA for TE's used as current converters. However, if the TE is used as a voltage converter,

or as a TVC in parallel with a shunt, the additional impedance is added directly to the relatively low heater re- sistance. The effect is large down to at least 20 mA and

significant to 5 or 10 mA. At 2.5 mA, where the heater re- sistance (about 400 S2) is high relative to that of the leads,

the ac-dc difference at 50 kHz is usually less than 5 ppm.

The skin effect problem was overcome in these TVCs

by using 5 mA and 10 mA TE's with platinum input wires.

They were made to NBS specifications by three cooperating manufacturers. Fortunately, platinum has an ex- pansion coefficient close to that of glass and makes a reli- able seal. Although the Peltier coefficient is large, the

heating effect is small at these low currents. The voltage converters designated FG and FH agree with NBS stan- dards to 5 ppm or better both as voltage and current converters. The differences are equally small down to well

below 20 Hz, and the corrections to a TCC are of course

quite independent of the shunt at low frequency [1]. TCC Construction, Connections, and

Grounding

The thermoelements are placed in insulating styrofoam

containers and mounted in short brass tubes. The input

is a GR874 coaxial connector and the thermocouple output

is through a two-contact connector which has very low

thermal EMF's at the contacts (Fig. 1). The 0.3-V TVC, designated "FG," is a 10-mA 30-fi TE

with no additional resistors. It also serves as a 10-mA

TCC—the lowest of the 14 ranges. The other TVC ("FH")

contains a 5-mA TE, with a heater resistance of about 90

fi, and a resistor in series. The combination is paralleled by a second resistor of 39 il so that FH becomes a 0.6-V TVC and a 20-mA TCC.

The three high current shunts are made of bifilar strips

of Ni-Cr quaternary alloy, insulated with 1-mil "Mylar"

and mounted between aluminum heat sinks (Fig. 1). The

shunt resistance material is gold plated at the ends, then

tinned and soldered to heavy copper end pieces to ensure good current distribution. The potential taps are near the centers of the shunt strip and are brought out at right angles to the shunt by a coaxial conductor passing through one half of the heat sink. A flexible cable extends the out- i put to a GR874 connector to which the TVC is attached.

The outside of the coaxial conductor is connected to one

half of the shunt, and the center conductor passes through \ 182 i
WILLIAMS: THERMAL CURRENT CONVERTERS 521

a hole in that part of the shunt and the insulation to the

other half where it is silver soldered.

The 10-20-A shunt resistor is 5 mils thick, 10 cm wide

and about 34 cm between potential taps. After being folded

into bifilar form the overall length is about 24 cm. The

resistor for the 3-6-A shunt is 2 mils thick and approximately 4 by 14 cm between potential taps. The 1-2-A shunt

is of the same material and approximately 1.8 by 25 cm.

Building the three shunts for the milliampere ranges

with available parts or materials was an awkward problem.

The required resistances were too high for bifilar strip

material and yet lower than the commercial resistors or- dinarily available. Tin-oxide resistors in parallel provided

a satisfactory solution. The 300-600-mA shunt (1 Q) contains ten 10-fi resistors, and the 100-200-mA range (3.33

Q) contains six 20-fl resistors. Parallel combinations of two

30-Q resistors and four 60-fi resistors were tried for the

30-60-mA shunt, but one 15-Q resistor was used as it had

the lower ac-dc difference. The resistors are mounted in 2-in diameter brass cylinders and connected between the

outer casing and a wire which joins the center contacts of

both the input and output connectors. (The input connector is out of view in Fig. 1.) Where more than one re- sistor is used they are arranged concentrically around the

center conductor.

When two shunts are connected in series for intercomparison tests care should be taken to avoid capacitive currents that might flow in parallel with one shunt after flowing through the other. Therefore a shunt casing (or

heat sink) should not be at the potential of the interconnecting conductor as in Fig. 2. It is recommended that one

casing be grounded and the other connected to the high (or

ungrounded) input conductor.

The heat sinks for the bifilar shunts should be connected

similarly. The connections shown in Fig. 3 is recom- mended, and it should not make an appreciable difference

whether one shunt is grounded or the other. The change

in ac-dc difference caused by changing the ground was

about 10 ppm at 50 kHz on the higher ranges of these

shunts. The effect was well under 10 ppm on lower ranges

and at 20 kHz.

TCC Calibrations

Several procedures may be used for ac-dc difference

measurements as described briefly in [3, section 6]. However, the TE comparator and the procedures described in

[3, section 7] were used for testing these TCC's. The TE

comparator affords a significant advantage where the

power supplies are even slightly unstable. The precision

of these tests probably could not be obtained without it. A fraction of the output emf of one TE is balanced against

the emf of the second to obtain an approximate null indication on a detector. Small changes in the ratio of emfs

resulting from ac-dc differences may then be measured by

changes in the detector indication. The detector may be

a microvoltmeter, although for these tests a galvanometer was used whose scale factor (ppm/mm) was determined

for each ac-dc difference determination.

Shunt

Casings

Fig. 2. TCC's in series.

Heat Sinks

Fig. 3. TCC's in series.

TCC's (shunt and TVC combinations) are usually cali- brated and used as one instrument with one correction for each frequency of interest. However the shunt and TVC

corrections can be separated for all but the lowest ranges

if measurements are made of the corrections to the TVCs

as voltage converters. It is necessary to know these cor- rections if ranges are changed by changing the TVCs.

Separating the shunt and TVC corrections has another

advantage. Several TVCs may be used, and if one is acci- dentally burned out the shunts may be used with others.A third TVC, designated "FE," was assembled using an

ordinary 20-mA TE to demonstrate the accuracy with

which TVC corrections may be applied. Its correction is +22 ppm at 50 kHz and is ascribed to skin effect in the

heater lead-wires. Shunt tests and intercomparisons with

TVCs FH and FE are discussed below.

The correction to the shunt and TVC combination (5c)

is not exactly equal to the sum of the shunt correction (6s)

and the TVC correction {5t). Separating these corrections

is complicated because the proportion of current carried

by the TVC varies with range. Hermach has developed a simple approximate formula which is accurate enough at currents of 100 mA or more

dc = m{5s + 5r) (1) where m = {I - It)/I, I is the total current, and It is the

current through the TVC. The multiplier m differs from

unity by only 10 percent at 100 and 200 mA, and its effect on 6c diminishes rapidly at higher currents. The formula

is more complicated at 30 and 60 mA and involves small

reactive components of the impedance which are extremely

difficult to measure. Corrections to these ranges are, therefore, determined by comparison with adjacent lower

and higher ranges as explained below.

In the intercomparison tests, measurements are made

183
522 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, DECEMBER 1976

TABLE II Intercomparisons of TCC's (ppm)

TCC^ TCCj Applied

Current

Amperes D D

FG FH 0.01 Q Q FG30 FH 0.02 + 1 + 11 FGIOO FH60 0.06 +10 +23

FG300 FH200 0.2 + 7 +20

FGl FH600 0.6 - 7 -33

FG3 FH2 2 +12 +22 FGIO FH6 6 -24 -54

TABLE III AC-DC Difference of TCC's (ppm)

TCC Applied

Current Amperes

20 kHz 50 kHz

6 measured computed measured

6 computed

FG 0.01 - 1 0 0 - 2

FH 0.02 0 0 - 2 - 2

FG30 0.03 + 3 + 1 +10 + 9 FH60 0.06 + 3 0 +12 +14

FGIOO 0.1 + 9 +10 + 32 +37

FH200 0.2 + 8 +10 + 36 + 37

FG300 0.3 +13 +17 +51 +5 7 FH600 0.6 +19 +17 +53 +57

FGl 1 +10 +10 + 24 +24

FH2 2 +12 + 10 +28 +24

FG3 3 +21 +22 +44 +46

FH6 6 +18 +22 + 36 +46

FGIO 10 + 8 - 2 + 3 - 8 FH20 16 + 5 - 2 + 2 - 8

of the difference between two shunt-TVC combinations.

If this difference is called "D" we may write

D = dci- dc2 (2) which becomes, after substituting from (1),

m^dsi — rn2ds2 = D + {1712^x2 -~ midri)- (3) At 100 and 200 mA the effect of the multiplier m (= 0.9)

is less than 5 ppm for the corrections to these shunts and

TVC's. Since there are other uncertainties of this magnitude the effect has been neglected. However, where the

corrections are 50 ppm or more it should be taken into account. A 7-step intercomparison was made at 20 and 50 kHz,

and the relative ac-dc differences are listed under D in Table II. Measurement of ac-dc difference corrections for

all ranges were also made at the same frequencies by

comparison with NBS standard current converters most

of which are specially made TE's. These corrections are

listed under "5 measured" in Table III. In two additional

columns in Table III under "6 computed" these corrections are determined from the intercomparison tests. The base ranges serving as starting points for the computed 6's were chosen arbitrarily. The "measured" cor- rections to the 20-mA range and the 1-A range were taken

as base points and the others were computed by using (2) and the data from Table II [2]. As mentioned above, the

TABLE IV

Intercomparison of TCC's with TVC Corrections (ppm)

Applied 50 kHz TCC Current

Amperes D SI S2 FG FE 0.01 + 3 FG30 FE 0.02 + 9 FGIOO FE60 0.06 +14 FG300 FE200 0.2 + 8 + 30 FGl FE500 0.6 -52 -30 FG3 FE2 2 0 +22 FGIO FE6 6 -54 -42

ACTABLE VDC Difference of TCC's with TVC Corrections (ppm)

Applied

Current

Amperes

50 kHz

TCC

*C measured

's computed computed

0.01 0 0 - 5

FE 0.02 - 8 - 8 - 8

FG30 FE60 0.03

0.06

+10 +18+ 1 +10

FGIOO FE2000.1

0.2

+ 32 +48

0 +32

+22 +26 +24+24

FG300

FE600

0. 3 0.6 +42 + 720 +42

+22 +50

+54

+ 54

FE2

1

2 +28+53

0 +20

+22 +31

+24

+24

FG3

FE6

36 +44

+6 3

0 +44 +22 +41

+46 +46

FGIO

FE20

10

16 + 3 +13

0 +3

+22 - 9 + 4 + 4

relationship of the shunt and TVC corrections is complicated at 30 and 60>nA where the TVC's carry 33 percent

of the current. Therefore, the 20-mA TCC (FH) is used as a starting point for computing corrections to FG30 and

FG.

Similar data are listed in Tables IV and V for 50 kHz,

but TVC corrections are applied in these tables since FE

is used. The corresponding ac-dc differences for 20 kHz

were smaller, as in Tables II and III, and are not shown.

Intercomparison differences are listed under "D" in Table

IV, and TVC corrections from (3), with mj and equal

to 1, are applied. (5^2 ~ ^ri = +22 ppm). The remaining

shunt differences are listed under "dsi — ds2- " At 60-mA,

equation (3) is not sufficiently exact, and, therefore, cor- rections to the 60-mA shunt and FE are not separated.

Corrections to each TCC with FE were determined rel- ative to NBS standards, and are listed under "6c" in Table

V. The corresponding values for the same shunt with FG

are taken from Table III. After applying the TVC correc- tions (dr = 0 for FG and +22 ppm for FE) the remaining

shunt corrections are listed under "6 measured."

Again, as in Table III, shunt corrections are computed

in Table V for all ranges beginning, as before, with the

measured correction to the 1-A and 20-mA ranges. The

computed correction to FE60 and FG30 apply to the

combination since the correction dc cannot be reliably

separated at this level.

184
WILLIAMS: THERMAL CURRENT CONVERTERS 523

The 20-mA range and the 1- and 10- A ranges are rec- ommended as minimum reference points to be tested rel- ative to other standards outside the set. Intercomparison

determinations would be based on the 20-mA and 1-A

corrections, and the 10-A range would serve to verify the

accuracy of the step-up tests for the highest shunt ranges. TVC corrections would also have to be determined.

The close similarity of measured corrections for the same

shunt at different current levels shows that the ac-dc

differences are not appreciably affected by the current. This similarity in Table III also shows that the TVC's FG

and FH, which are changed when the current is changed,

have nearly the same correction. It is very unlikely that one change would offset the other on so many ranges. The

similarity of measured and computed corrections in Tables

III and V shows that the intercomparison tests can be

made quite accurately.

Using Thermal Current Converters

Ac-dc difference measurements may be made of transfer instruments and ammeters which respond nearly equally

to ac and dc currents. The TE comparator and procedures

referred to earlier may be used if the test instrument has

a thermocouple output voltage which can be connected to the comparator. Otherwise an indicator or readout in- strument is observed which is usually a part of the device

under test.

If the test instrument does not respond to direct current

it may be tested by two or three procedures, but the one

outlined below is recommended. Changes in detector in- dication are observed to determine the difference between

the ac current and an accurately measured dc current nominally equal to it. The suggested circuit is shown in Fig. 4, and the order

of currents applied to the standard TCC is indicated

(DC-I-, AC, DC—). The potentiometer P is adjusted for an

approximate null indication on the detector D. Detector

readings are then made for each dc current after it has been

adjusted to the nominal test value. The ac reading Ea is made after the ac has been adjusted for the desired indication of the test instrument. The two readings on dc are averaged (Ed), and the ac may then be calculated:

where /dc is the dc current, E is the TE EMF indicated by

DC+ AC DCFig. 4. Circuits for testing an ac ammeter relative to dc standards.

the potentiometer P, 5s is the ac-dc difference of the

standard TCC and n = AE/AI. The factor n is a measure

of the deviation of the TE from a square-law response and

is usually 1.7 to 1.9 at rated TE current, (see [1, section

9]).

Conclusion

It has been shown that a TCC set consisting of six shunts

and two TVC's can be constructed to cover fourteen ranges from 10 mA to 20 A. Intercomparison tests can be made in seven steps to assist in maintaining accurate ac-dc dif- ference corrections. Thermoelements with platinum heater

lead-wires are preferable, because the TVC corrections are

negligible. However if ordinary TE's are used the TVC

corrections can be accurately determined and applied. The

estimated accuracies from 20 Hz to 50 kHz are 50 ppm at 6 A and lower and 100 ppm at 10 and 20 A. Measurements

of ac current would require additional corrections for the

dc standards used as reference and considerations of the

precision of the instrument under test. Acknowledgment

This work was supported in part by the Army Metrology

and Calibration Center, Redstone, AL. Their encouragement and support are gratefully acknowledged.

References

(1) F. L. Hermach, "Thermal converters as ac-dc transfer standards for current and voltage measurements at audio frequencies," J. Res. Nat.

Bur. Stand., vol. 48, pp. 121-138, Feb. 1952.

[2] F. C. Widdis, "The theory of Peltier and Thomson effects in thermal ac-dc transfer devices," Proc. Inst. Elec. Eng., monogr. 497M, Jan.

1962.

[3] E. S. Williams, "Thermal voltage converters and comparators for very accurate ac voltage measurements," J. Res. Nat. Bur. Stand., vol. 7,5C, nos. 3 and 4, pp. 145-154, July-Dec. 1971.

[4) F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency voltage measurements of high accuracy," IEEE Trans. Instrum. Meas., vol. IM-15, pp. 260-268, Dec. 1966.

185
290 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 198S

Section 13

A Dual-Channel Automated Comparator for AC-DC

Difference Measurements

EARL S. WILLIAMS and JOSEPH R. KINARD, senior member , ieee

Abstract—\n automated ac-dc difference calibration system is described. The system incorporates a new electronic comparator which

determines ac-dc differences of thermal voltage converters (TVC's)

by simultaneously measuring the difference between nearly equal ac and

dc voltages with both the test and standard instruments. The comparator consists essentially of two practically identical channels each containing a digital-to-analog converter balancing circuit, an operational

amplifier to amplify voltage imbalance, and an integrator circuit in which

a capacitor is charged during an accurately controlled 10-s period. The

difference between the ac and dc voltages applied to the test and stan- dard TVC's is computed from differences in capacitor voltages, and the ac-dc difference is derived from the variation between the test and stan- dard channel indications of this voltage difference. Measurements are made in about half the time required for our manual procedures.

I. Introduction

AC AND DC voltages are compared using a thermal voltage .converter (TVC) which responds nearly equally to acand

dc voltages. Those discussed here are electrothermic. Typically,

they consist of ranging resistors and a thermoelement— a vac- uum bulb containing a heater wire with a thermocouple at- tached to its midpoint. The output, E, of the thermocouple

is usually 7-12 mV with rated heater current. The TVC is

Manuscript received August 23, 1984. This work was supported by

tbe Calibration Coordination Group of the U.S. Department of Defense.

J. R. Kinard is with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

E. S. Williams (retired) was with the Electricity Division, National Bureau of Standards, Gaithersburg, MD 20899.

connected alternately to an ac voltage, V^, and to both polarities of a dc reference voltage, V^j. is measured relative to by observing the output EMF, E, of the TVC and changes

in output, Ea-Efj, where E^ is observed with applied, and

E^ is observed with Vf^ applied. When the difference between

and V(j is small, the proportional difference can be computed from (1):

{Vg - Vd) _ {Eg- Eg) ^ ^ Va nE

The factor n relates small changes in TVC heater current to corresponding changes in output EMF, E. It is near 2.0 (a square-law response) at very low heater currents, but is usually 1.7-1.9 at rated heater current. 6 is the ac-dc difference of the TVC. It is negligible at low frequencies, but becomes

significant at frequencies of 20 kHz and higher in most TVC's. The ac-dc difference correction for a TVC is determined by

comparing it to a similar standard instrument whose correction,

5^, is known [1]. Ac and dc voltages are applied to both TVC's

in parallel, and in most procedures, the ac and dc voltages are adjusted for equal outputs from the test TVC, i.e., Ea = E^

(1). The test instrument thus indicates that = F^. The ac- tual proportional difference between the ac and dc voltages is then measured by observing E and changes in E from the stan- dard TVC. The correction to the test TVC, 5,, is computed

using an equation similar to (1):

186
WILLIAMS AND KINARD: DUAL-CHANNEL AUTOMATED CONVERTER 291

IEEE 488

IEEE 488

DC SOURCE

DC DVM ~7K~

AC SOURCE

CONTROL FROM

IEEE 488 CH RELAY

AC DVM

F-REQ. / COUNTER VIEEE 488

DUAL CHANNEL COMPARATOR

8TD. TVC

VOLTAOE

NULLING DAC

IEEE 488

TEST TVC

VOLTAOE

NULLING DAC

INTEGRATOR

) DVM

INTEGRATOR

IEEE 488 CONTROLLER

PRINTER PLOTTER >

Fig. 1. Block diagram of automated ac -de difference calibration system.

(2)

where subscripts t and s refer to test and standard. With a manual comparator, the precise adjustments of ac and

dc voltages for equal outputs from the test TVC are tedious and

a skilled operator is necessary [2] . These adjustments have

been made automatically but the process is slowed significantly

[3] . In the alternative procedure used in the new comparator,

data is coOected more quickly because the ac and dc voltages

are set nearly equal and not further adjusted during one deter- mination. The ac voltage is trimmed automatically after each

determination to compensate for drift and to avoid the dif- ficulty, discussed below, of a large difference between ac and

dc appHed voltages. In this new approach the difference be- tween the unequal ac and dc voltages is measured by both the

test and standard TVC's, and, therefore.

iVa-V^) iE,,-E^,) {Eg, - E^,)

+ C)c- TTT. + 0f (.j) nsiEs) ntiEt)

Eliminating the first expression and solving the remaining two

for 5,, we have

Of :-: ~ + 0, ^siEs) ntiEt)

(4)

II. Automated System

The entire automated ac-dc difference calibration system is shown in block diagram form in Fig. 1. A photograph of the equipment appears in Fig. 2. The system contains programmable ac and dc voltage sources, a high-voltage relay for switching between ac and dc, voltmeters and frequency counters, a desktop computer to provide IEEE-488 bus control, and the two-channel comparator itself. The two-channel comparator

circuit is sketched in Fig. 3 . It uses elements from a commercial

data acquisition system (DAS) including programmable digitalto-analog converters (DAC's), programmable relays (Kl to K6

and K9), and an auto-ranging voltmeter (VM). An additional board was assembled containing the DAC balancing circuits, amplifiers Al to A4 and their related resistors and capacitors.

The DAC's supply up to 10 volts to Lindeck-type circuits with S-kfi total circuit resistance. With only one range, the

12-bit resolution is insufficient for an appropriate null. A twopole four-position input switch allows the operator to select either a 5- or lO-.Q resistance in the divider circuit, thus pro- viding either a 10- or 20-mV range. The measurement procedure is as follows. The operator provides ranges, voltages, frequencies, and record-keeping information to the controller interactively. The ac and dc supplies are then programmed to the appropriate values, and dc is applied

to the TVC's. With the DAC set to zero and K3 closed, the standard TVC output, E^j^, is measured by the voltmeter, and

a DAC voltage is computed and supplied to null the input to amplifier Al . The output of Al , with gain of 10 000, is then

measured by closing relay K2, and the DAC signal is trimmed

in several steps until the amplifier output is less than 20 mV.

The same balancing procedure is followed in the test channel. When both balances have been completed, «j(£'j) and HjiEj)

are computed and printed along with other test parameters.

187
292 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

Fig. 2. Automated ac-dc difference calibration system.

K3^

Fig. 3. Dual-channel ac-dc difference comparator.

Voltages are then applied to both TVC's in the test sequence

(ac, dc+, dc-, ac), and after each one has stabilized, the relay

K7, with a contact in each channel, is closed to charge the two

capacitors during an accurately measured time interval of 10 s. The capacitor voltages, four from each channel for a single

measurement, are then measured with the voltmeter by closing

relays Kl and K4 in turn, after which the capacitors are dis- charged by closing two-pole relay K8. Providing they are stable over the time required for one determination, amplifier offsets cancel because the results are collected in paired sums and

differences. For the standard channel, the relation between the voltage,

ejn, at the input to Al and the voltage, eQ^^^,on the integrator

capacitor is

1 r Rl + R2 j'-^^^ndr = -e,^XlO^ (5)

with Rl = 10n,R2 = 100kn,R3 = 1 Mn,Cl = 1 ^F,^nd an

integration time of 10 s. Nominally equal values of resistance and capacitance are used in the standard and test channels, so the same nominal relations apply. If the eight voltage readings

are reversed in sign and set equal to ^2,^4,^6, and in the standard channel, and e^, e^, e^, and eg in the test channel,

then using (5), (4) may be written as

5 X (^2 - ^4 - ^6 + ^8) 5 X (eg - gg - + g9) , , 0,= Z 2 +0r.

(6)

The numerator in (6) gives twice the actual change in EMF, so the constants are adjusted appropriately. These constants are

also adjusted slightly to compensate for departures from nominal in the integrators and amplifiers. The test TVC correction

is computed using (6), with the appropriate ac-dc difference

correction, 5^, determined from an equation devised to simulate

the standard TVC correction curve. Since this two-channel method requires the knowledge of

n for both test and standard, any uncertainty in either n contributes to uncertainty in the final ac-dc difference. Should

either or both terms in (4) become large due to inequality be- tween applied ac and dc voltages, the calculation contains the

difference of two large terms each involving an n. Since the n

value can only be determined to 1 or 2 percent, a constraint is added to system that the 1st term in (4) be 50 ppm or less to

limit the uncertainty contributed by n to 1 ppm, or less. With

this constraint the procedure places an upper limit of about 50

ppm on the difference between the apphed ac and dc voltages.

Equally satisfactory results are obtained by a "two-voltmeter"

188
WILLIAMS AND KINARD: DUAL-CHANNPX AUTOMATED CONVI'.RTER 293

1008V

100V

18V

18 100 Ik 10k FREQUENCY (Hi)180k IM

Fig. 4. Voltage and frequency limits of calibration system.

method similar to that used by Inghs in a semi-automatic sys- tem [4] . Instead of the analog integrators, an external digital voltmeter connected to relay K9, and the voltmeter located in the DAS, are used to measure the amphfied voltage imbalances.

After the DAC balances are made, relays K2 and K9 are closed and remain closed while the ac and dc voltages are applied. The

voltmeters are read alternately and the two sums are accumulated. If ten readings are summed for each channel with the measurement sequence ac, dc+, dc-, ac, and the sums set equal to e2 through and through e^, then 6, can still be

obtained from (6) with no change.

III. Other Measurements

Measurements of the response characteristic, n, and comparisons of ac and dc voltages are two types of measurements made

using only one channel of the comparator. For these cases, each EMF measurement is the average of 10 voltmeter readings

taken in about 10 s.

A. Measurements of n The factor n is ordinarily measured at 10 percent intervals from 50 to 110 percent of rated vohage. For instruments to be calibrated at rated vohage, however, // is usually determined

at 80, 90, and 100 percent. In the middle voltage range, small accurate voltage steps are easily made and the stability of the system is optimum. Tests are made, therefore, most conveniently on the 100-V range of multirange TVC's, and the re- sulting n values are usable with any range when related to the TVC output, E. Low-voltage single-range TVC's are usually

tested with additional resistance in series, so that 50-100 V

may be applied. The output of the TVC is connected to the "STD" channel, and the operator enters the lowest and highest

voltages applied. After the TVC has stabilized at the measure- ment voltage, the DAC balance is made as described above, and

relay K2 is closed connecting the amplifier to the voltmeter. The TVC input is increased and decreased alternately by 0.1 V,

AV, and the resulting changes in TVC output are computed

from voltmeter readings.

With only dc applied, the relationship between small changes

in input voltage, AV, and changes in the output voltage, AE

,

can be given by rewriting (1) as AV/V^AE/nE or n ^ {AE/E)liAV/V). (7)

As in the two-voltmeter method, 10 voltmeter readings of the output are averaged to obtain one measured value. One AE

is calculated from measurements taken in the sequence {V,

V + AV, V). The last reading with V applied from the previous

set is then taken as the first reading in the next set. Four values of n are computed from four values of A£" requiring nine volt- meter measurements. These data are then used to determine

constants for a straight hne from which the appropriate value of n can then be found for any value of E.

B. Comparison ofAC and DC Voltages

AC voltage, such as from an ac cahbrator, can be measured

relative to a dc -reference voltage by a procedure similar to that used for the determination of n. The system applies dc+, ac, dc-, to a TVC in that order, and its output is monitored by

the "STD" channel with the DAC balance and voltmeter readings performed as described above. Computations are made

using (1 ) with {E^ - computed from three vohage readingsej, ^2, ^3—so that

2{Ea- Ea)-e,+2e2 -

where ei and e^ correspond to the two directions of dc volt- age, and ^2 to the ac voltage. Again, four values of EMF dif- ference are computed from nine voltages— each being the aver- age of 10 vohmeter readings. When low voltages are being

compared, the voltage drops along the leads from the sources

to the TVC may become significant. Dedicated cables are provide with total inner and outer resistance equal for both ac and

dc. These connections also permit voltage sensing of the dc if required.

IV. Results

A significant reduction in the time required to perform routine cahbrations is achieved with this system. Four determinations, for one set of test conditions, are made and averaged in about half the time required with our manual procedures. The

voltage and frequency operating ranges, shown in Fig. 4, are determined primarily by the availability and cost of a program- mable ac voltage source. The region between 100 kHz and 1 MHz represents a maximum volt-Hertz product of 10^. Test

189
294 IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. IM-34, NO. 2, JUNE 1985

results taken over the operational region rarely differ by more

than 10 ppm from data obtained manually. The standard devia- tion is generally less than 5 ppm in the middle -voltage and middle-frequency region indicated approximately by the cross- hatched region in Fig. 4. For well-constructed single-range

TVC's, standard deviations of 2 ppm or less are achieved. At extreme voltages and frequencies standard deviations are usually less than 10 ppm. Although this system was built as a re- search tool for investigation of new thermal voltage conversion

techniques and instruments, it has been used for some routine

calibrations. Another system, of very similar design and dedicated to routine calibration work, is under construction. These

automatic systems are expected to fulfill most of our TVC cali- bration requirements.

Acknowledgment

The authors would like to thank F. L. Hermach for his helpful suggestions.

References

[1] F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency measurements of high accuracy," /£'£'£' Trans. Instrum.

Meas., vol. IM-15. pp. 260-268, Dec. 1966.

[2| E. S. Williams, "Thermal voltage converters and comparator for very accurate ac voltage measurements," /. Res. Nat. Bur. Stand.,

vol. 75C,pp. 145-154, 1971.

[3] , "A thermoelement comparator for automatic ac-dc difference measurements," IEEE Trans. Instrum. Meas., vol. lM-29, pp. 405-

409, Dec. 1980.

[4j B. D. Ingbs, "Evaluation of ac-dc transfer errors for thermal converter-multiplier combinations," Metrologia, vol. 16, pp. 177-

181, 1980.

190
Section 14

GENERAL INSTRUCTIONS FOR THE OPERATION OF NBS AUTOMATED

AC-DC COMPARATOR SYSTEM II

Thomas E. Lipe

Electricity Division

National Bureau of Standards

December, 1987

191
INSTRUCTIONS FOR THE SETUP AND USE OF

NBS AUTOMATED AC-DC COMPARATOR SYSTEM II

General Description

The new NBS automated ac-dc comparator (System II) was

built in 1985-1986 to assist in the everyday calibration

services of the group. It is expected that this system

will eventually be responsible for the majority of the

ac-dc calibrations performed in the Electricity

Division.

The new system is composed of two racks of calibration

equipment, a desktop computer which acts as the system

controller, and several input/output devices. The racks

are divided into an ac rack and a dc rack as follows:

From the bottom, the ac rack contains:

a) Relay box and switch panel built in-house

b) Ac voltage source (Fluke 5200A)

c) Frequency counter (Systron-Donner 604 lA)

d) Ac voltmeter (Fluke 8922A) and IEEE-488 translator

(Fluke 1120A)

e) Ac high-voltage power amplifier (Fluke 5215A)

The dc rack contains:

a) Dc voltage source (Fluke 5440B)

b) Data acquisition system (Hewlett-Packard 3497A)

c) Dc voltmeter (Fluke 8840A) and IEEE-488 bus optical

isolator (ICS 4830B)

d) Dc nanovoltmeter (s) (one or two Keithley 181

nanovoltmeters)

System control and support:

a) System contoller (Hewlett-Packard Series 200/300

desktop computer)

b) Printer (Dataproducts Prism P-132 matrix printer)

c) Plotter (HewlettPackard 7475A six pen plotter)

d) Access to administrative minicomputer (DEC VAX

11/750)

e) Administrative printer (Hewlett-Packard LaserJet+

laser printer)

Future expansion:

a) Hewlett-Packard 3498A Expander Box for multi-channel

calibrations

192
Note that the product names in parentheses indicate the

equipment supported by the system software as of August

27, 1987. Future versions of the software will have the

capability to support a variety of equipment.

3. Although most of the equipment in the system was

procured from commercial sources, the "guts" of the

system (the relay box and several circuit boards in the

data acquisition system) were built in-house. The relay

box and its associated switch panel contain two sets of

relays used for switching the output between ac to dc.

The designation of the relays to be used depends upon

the applied voltage with the crossover point at 70 volts

to coincide with the use of the high voltage interlock

system. The high voltage relays are Goto 8407 SPST

relays and the low voltage relays are Potter and

Brumfield KRP14DG 3PDT relays which have been modified

to a single throw configuration. By using two different

sets of two relays, problems such as arcing on high

voltage and contact resistance on low voltage are

avoided. Since there is no voltage on the output of the

relays in the normally open position, there is a built-in safety factor; when the relays are open, no

voltage is applied to the output connector. The

function of the front panel switches are as follows:

i) OUTPUT GND-OFF-ON: This switch controls the voltage

output to the units under test. In the 'GND'

position, the output of the system is connected to

the system ground at the ac supply. In the 'OFF'

position, the output is open circuited, and in the

'ON' position, the output is connected to the units

under calibration. This switch provides a second-stage safety measure, since the voltage

output can be immediately disconnected from the test

units, and because this switch must be activated in

concert with the output on/off relay,

ii) DETECTOR SWITCH 1-2-3-4: Because the DACs which

null the output EMF of the units under test have an

upper limit of 10.2375 volts, they may occasionally

be overranged. To prevent this, a voltage divider

(in the data acquisition system) is wired in series

with the output EMF. The input to the divider is

controlled by this switch, and its position is

determined by the output EMF of the units under

test. You will be prompted by the software to set

the switch to the proper position during a calibration.

iii) AC INPUT SELECT HI-OFF-LOW: When the Fluke

5200/5215 ac calibrator/ amplifier is used, the ac

voltage enters the relay box along separate leads

for high and low voltage. This switch selects the

high or low ac inputs and is set by the user before

the calibration process begins. On the high

setting, this switch also introduces a high-voltage

oscilloscope probe in series with the ac voltmeter

193
so as not to overrange it.

iv) MODE SELECT AUTO-DC-AC: This switch controls the

ac-dc relays and determines whether the relays are

under manual or computer control. In the 'AUTO'

position, both sets of ac-dc relays are controlled

by the data acquisition system. When switched to

'DC, the dc high-voltage output relay is closed

(output on) , providing the dc output to the output

connector. On 'AC the ac relay is closed (output

on) , providing ac voltage to the output connector.

Note that, in the present configuration, the high

voltage relays are the only ones controllable from

the front panel. This will be modified in future

versions.

V) CONNECTIONS TO THE RELAY BOX: The connections on

the front panel, from left to right are: the test

and standard channel detector leads and the voltage

output coaxial connector, and the ac low voltage

input and sense coaxial connectors. On the back,

from left to right, are the ac line connector for

the relay pov/er supply, the ac high voltage

connector, the ac voltmeter coaxial connector and

seven-pin control connector from the data

acquisition system, the detector cable to the data

acquisition system, the twin-pin connector for the

high voltage shield cable, and the dc voltage input

and sense connectors. The dc voltmeter lead may be

plugged into either the dc input or sense connector,

and a ground strap may be run from the dc input low

side to the ground terminal on the ac supply.

Initial Equipment Setup

4. Turn on all the equipment in the two racks. The ac line

switches for this equipment are on the front panels of

the individual units. Exceptions to this instruction

are the ac and dc calibrators, which should remain in

standby mode instead of being shut off after use (to

keep them warm) , and the computer, which is turned on at

the initial program load.

5. For the initial equipment setup, connect the test and

standard units in parallel through a coaxial tee

structure, with the voltage input at the free leg of the

tee. There are several variations in the connection

scheme, most notably the inclusion of a protection

device in series with the units under test, and a

voltage dropping resistor, also in series with the test

units. The first case usually takes the form of a pair

of back-to-back Zener diodes in parallel with the units

under test and an expendable thermcal element in series

with the combination. The diode voltages are chosen to

be about twice the applied voltage. When a large

194
overvoltage occurs, the diodes conduct, blowing out the

thermal element serving as a fuse, and protecting the

test units from being overranged. In the case where the

calibration voltage is less than ten volts, a series

voltage-dropping resistor may be added to the circuit to

force the supplies to run at a higher voltage range

where they are generally more stable. The voltage drop

due to the series resistor and/or the protection is

accounted for in the software.

6. Connect the detector leads to the output connectors on

the thermal elements. The detector connectors are

polarized to accommodate NBS standards which, naturally,

are of the opposite polarity to practically everyone

else's. In the future, this polarity difference will be

corrected for in the software; for the present, a polarity changing cable needs to be inserted in series

between the TE output and the detector lead.

7. Set the switches on the relay box front panel.

Initially, the output on/off switch should be set to

'OFF' (until the voltage is applied) , the detector

switch is irrelevent, the ac selection switch should be

set to either 'HI' or ' LOW , depending on the applied

voltage, and the mode switch set to 'AUTO'. If the

Fluke 5215A high voltage power amplifier will be used,

disconnect the input and sense leads from the Fluke

5200A ac source at the front of the relay panel. This

will eliminate potentially troublsome grounding

problems

.

8. For test voltages of 60 volts or greater, the high

voltage safety shield must be set up. Since the shield

is interlocked to the high voltage relays (always used

at such voltages) , failure to set up the shield will

prevent the relays from closing. To install the shield,

simply align the cutout in the end of the shield with a

matching cutout in the side unit and press the coaxial

phone plugs into their respective jacks.

Initial Computer Setup

9. Like most desktop computers, the Series 200 and 300

computers must be booted (the operating system must be

loaded) from disc before use. The initial program

loading utility for the new automated system includes

both the bootable system and the ac-dc calibration

supervisor, so that all the necessary software is loaded

in the beginning of a session without operator

intervention. The software is designed with the idea

that, once in the calibration supervisor program, the

operator need not exit the package to perform any

function. This is not only a more elegant approach than

having to load all the utility programs as needed, but a

195
much more efficient method when confronted with the

outrageously slow access time of the Model 82901M disc

drive used on the 9816 computer system.

Note: The following instructions assume that the

software being used is on either 5.25 inch or 3.5 inch

flexible discs. The operating system and supervisory

routines may also be on a 40 Mbyte Winchester hard disc

system. In the latter case, the system will boot from

the Winchester drive and no flexible discs will be used.

10. To load the ac-dc package, insert the flexible disc

marked SYSTEM in drive 0 and the disc marked BINARY FILE

LOADER in drive 1. On the 983 6 computer drive 0 is the

internal drive on the right hand side, drive 1 is the

left; the 9816 uses external drives (models 82901M and

9122D) which have the opposite convention (nothing like

standard ization!). After inserting the discs, turn on

the computer system. For the 983 6, the power switch is

at the lower right edge of the cabinet; on the 9816, the

power for either disc drive is on the front panel, and

the power switch for the computer is in the back of the

cabinet. The power switches for the Series 3 00

computers (model 310) is on the left front of the CPU

box and the left front of the monitor.

11. Upon powerup, the computer will load the BASIC 5.0

operating system, some necessary binary programs and

data files, and the ac-dc calibration supervisor. If

the date and time have not been set, then you will be

queried to enter the correct date and time. At the

completion of the boot, a command option menu will be

displayed with which the user can select the operation

of interest. These option are as follows:

SETUP I - INSTRUMENT CALIBRATION MENU

le;K0: FLUKE 54 OB: Allows calibration of Fluke

54 OB multirange transfer standards. le;Kl: HOLT

MODEL 11: Allows calibration of Holt Model 11

coaxial transfer standards,

i) K2: HOLT MODEL 6A: Allows calibration of Holt

Model 6A multirange transfer standards,

ii) K3: GUILDLINE 7100: Allows calibration of

Guildline 7100A multijunction transfer standards,

iii) K4: BALLANTINE 1600: Allows calibration of

Ballantine 1600A multirange transfer standards,

iv) K5: BALLANTINE 1605: Allows calibration of

Ballantine 1605A automatic multirange transfer

standards.

V) K6: NBS COAXIAL TYPE: Allows calibration of

coaxial single-range units similar to the NBS F7

set, such as the Julie TNB-102, Ballantine 1398A,

and old Hermach-Engelhard transfer sets.

196
viii) K9: MENU ->: Go to setup menu II. For the

instrument calibration menu listed above, pressing

the SHIFT key down simultaneously with the desired

function key will cause the system to calibrate an

instrument using a previously defined standard

calibration file. Pressing the function key alone

allows the operator to input all the necessary test

parameters

.

SETUP II - UTILITIES MENU

i) KO: GRAPH: Uses the series 200/300 graphics

capabilities to plot such relationships as the

thermal element response curve, the ac-dc difference

versus frequency of a thermal converter, etc.

ii) Kl: SYSTEM STATUS: Displays the current system

status

.

iii) K2: HELP: Accesses the ac-dc supervisory help

facility.

iv) K3: VAX: Boots the series 200/300 terminal

emulator package to allow data communication with

the Electricity Division's DEC VAX 11/750.

V) K4: STD CAL FILE: Permits the creation and

modification of standard calibration files.

vi) K5: STAND. CORR. : Creates and modifies the data

file containing the coefficients of the

characteristic curve for each of the NBS standards,

as well as their ac~dc difference corrections,

vii) K6: REPORT: Accesses the automated reporting

facility.

viii) K7: AMP. BOARD: Accesses a utility to measure the

amplifier gains and integration constants of the

amplifier board,

ix) K8: EXIT: Exits the ac-dc package and returns

processing to the BASIC 5.0 system command level,

x) K9: <- MENU: Go to setup menu I.

12. Upon pressing the appropriate function key, you will be

instructed to insert the proper supervisory disc in

drive 0, and, in the case of the instrument

calibrations, a data storage disc in drive 1. When

processing is continued, the system checks to see that

the proper disc is in drive 0, warns you if it's not,

and gives you another try. If the proper disc is

loaded, the system loads the appropriate supervisory

program and processing resumes with that subroutine.

All of the options in the setup menus, with the

exception of 'SYSTEM STATUS' are separate subroutines on

different discs, and are documented as such. For the

instructions on a particular option, please see the

appropriate section of this documentation. If the mass

storage device is a Winchester drive, the data file will

be placed on an appropriate volume (section of the

197
disc) , and the calibration master files will be read

from the system volume. In this case, you will not need

to change floppy discs.

At the end of a session, pressing the 'EXIT' key (either

setup menu II, K8) will return processing to the BASIC

5.0 system. If you want to run the ac-dc package again,

insert the BINARY FILE disc in drive 0, and press K5

('SCRATCH') on the system function key menu. Then press

K6 ('LOAD " ') and enter ACDCMSTR, followed by a double

quote. After the ac-dc supervisory program is loaded

(the light on the drive goes out and the drive quits

making noise) , press the RUN key. The master program

will run as described above, except that the date is

already known, and the session will proceed with the

initial setup menu. If the ac-dc master program is

still in memory, simply insert the BINARY FILE disc in

drive 0 and press the RUN key. Follow these guidlines

for systems with a Winchester drive, without, of course,

inserting floppy discs.

198
INSTRUCTIONS ON THE USE OF THE SYSTEM STATUS UTILITY

The system status utility (SYSTAT) may be called from

any of the supervisory routines in the ac-dc software

package. SYSTAT displays the present system status and

allows you to modify the characteristics of the

computer, serial port, and IEEE-488 bus.

When the package is first booted, SYSTAT is called and

accesses a data file containing the last definitions of

the IEEE-488 bus structure. When the status utility is

initiated by a function key interrupt, the utility

queries the computer system for information, and

displays this along with the data read in from the

IEEE-488 bus structure data file.

The status display is divided into three sections: the

computer status, IEEE-488 bus status, and serial

interface status as follows:

i) COMPUTER STATUS:

a) BASIC version: The currently loaded version of

Hewlett- Packard BASIC.

Available memory: The memory space currently

available to the program.

Mass storage device: The present default mass

storage device.

The present "PRINTER IS"

b)

c)

d)

e)

Printer interface:

device

.

Plotter interface:

device.

The present "PLOTTER IS"

ii) IEEE-488 BUS STATUS: Seven lines are available to

display the bus addressable instruments which the

system will recognize. These are displayed along

with their addresses.

ii) SERIAL STATUS:

a) Serial address: The internal address of the

serial interface in the series 200 computer.

b) Baud rate: The present transmit/receive speed

of the serial port.

In addition to the display, a function key menu is

displayed at the bottom of the screen. This menu allows

you to modify some of the characteristics of the system

as follows:

i) KO: MODIFY COMP. : This key allows you to change

the default mass storage device identifier, and the

present printer and plotter interface,

ii) Kl: MODIFY 488: Use this key to modify the

IEEE-488 bus structure. Up to seven instruments may

be defined in the system.

199
iii) K2: MODIFY SERIAL: Allows you to change the serial

interface select code and to modify the

transmit/receive speed,

iv) K3: WRITE TO FILE: Writes out the new IEEE-488 bus

information to the system status data file.

V) K8: SYSTAT EXIT: Returns processing to the calling

subroutine

.

5. When any of the modification keys are pressed, an arrow

(->) is displayed to the left of the section undergoing

modification, and the system directs you to select the

field to modify. This is done by pressing the SHIFT key

and rotating the knob, causing the arrow to move

vertically in the left hand column. When the arrow

points to the field to modify, pressing the ENTER key

will display a prompt into which you may type the

pertinent information. When the IEEE-488 structure is

modified, an additional prompt queries the operator for

the address of the unit under modification. Please note

that the instrument's function must be explicitly typed

out: for example, a Fluke 5440A dc calibrator must have

the string "DC CALIBRATOR" in the definition; the

definition for a Hewlett-Packard 3497A data acquisition

system must contain the string "DATA ACQUISITION

SYSTEM", although it may be in lowercase text. This

aids in defining variables used to output commands to

the various bus instruments during a calibration. Also

note that at the present time, no check is made to

verify that the instruments defined by SYSTAT are

actually connected to the bus. After the appropriate

field is modified, the updated status is displayed and

the function key menu returns. If the present IEEE-488

bus structure definitions need to be saved, pressing K3

(WRITE TO FILE) will save the present definitions in the

file SYSTATDAT. Pressing K8 (SYSTAT EXIT) will return

processing to the calling program.

6. When SYSTAT is called by a routine other than the ac-dc

program supervisor (for example, the n-test routine)

,

modifications are not permitted. The present system

status is displayed, and you may press CONTINUE to exit

SYSTAT and return to the calling program.

200
INSTRUCTIONS FOR USING THE AMPLIFIER GAIN CHECKING UTILITY

Introduction and setup

1. Using an automated ac-dc comparator utilizing amplifiers

and analog integrators entails knowing the amplifier

gain factors and the time constants of the analog

integrators. These quantities are usually measured by

supplying a small voltage to each channel of the

detector and working the relays in the Data Acquisition

System to connect the voltage to be measured to a DVM.

A stopwatch is used to clock the integration time of the

analog integrators. From this method, it is apparant

that many sources of error are present, from the

instability of the voltage supply to the precision of

the timing. The utility described here uses voltages

supplied by an adjustable DAC to measure the gains and

time constants, and the timing is controlled by a

computer. In this way the error associated with the

timing and sensitive adjustments to a potentiometer are

avoided.

2. To enter the amplifier verification utility, press

K7:'GAIN CHECK' in the master calibration supervisory

option menu. You will be instructed to insert the

Utilities disc in drive 1 and press 'CONTINUE' to load

the utility. If a Winchester drive is designated as the

mass storage device, the utilities will be read from the

system volume of the Winchester, and disc swapping is

unnecessary.

3. After the amplifier verification utility is loaded, you

will be instructed to short the inputs to the detector

leads. This is done to check the offsets in the

first-stage amplifiers and can be accomplished by

inserting a shorting plug into the input connector of

each detector lead. After the leads are shorted, press

'CONTINUE' to begin the verification routine.

Measurement of amplifier Board Characteristics

4. The verification routine begins by presenting a display

composed of three windows. These windows and their

functions are described below:

i) Message window: This window displays the current

action (checking offsets, discharge voltage, etc.)

being taken by the routine,

ii) Summary window: This displays a running summary of

the pertinent information taken by the computer,

iii) First-stage amplifier window: This window displays

data relating to the offsets and gains of the

first-stage amplifier in each channel,

iv) Integrator window: This window displays the

information relating to the constants associated

with the analog integrators. As data is taken and

201
reduced, the raw data and a summary of the data will

be displayed in turn.

5. Actual checking of the amplifier board begins by

measuring the offset of the first-stage amplifier in the

standard channel . This is associated with the

first-stage amplifiers and is generally about 0.5mV.

After getting the offset, the voltage needed to supply

ImV to the channel is calculated and applied to the DAC.

The routine then disconnects the input to the amplifier

from the DVM and connects the output of the first-stage

amplifier. The system pauses to let the DVM settle and

reads the output of the amplifier. The offset and

measured amplifier gains are written out to the screen

and the process is repeated twice more. After three

determinations of this data are taken, the amplifier

characteristics of the test channel are measured.

6. In order to check the analog integrators, a voltage of

IV is needed at the input to the integrators. The

standard channel DAC is adjusted until IV is measured at

the output of the first-stage amplifier. The discharge

relay is closed and the discharge voltage measured and

displayed on the screen. The discharge voltage tends to

be variable, but should be very small (tenths of

millivolts) . After reading the discharge voltage, the

discharge relay is opened and the output of the

first-stage amplifier connected to the integrators.

Integration is timed for ten seconds and the resulting

voltage read by the DVM. The integration is displayed

on the DVM so you can check the integration rate. The

integrator output and 1/RC time constant are displayed

on the screen, and two more determinations are taken.

The test channel is then measured in the same manner.

7. Note that, during these measurements, the status of the

system is displayed in the message window.

8. After the integrator characteristics are checked, the

constants for the n-test and calibration routines are

calculated and displayed in the summary window.

9. Upon completion of the measurements, the previous date

of board verification and the constants currently being

used are presented in the message window for comparison

with the new constants being displayed in the summary

window. A function key menu is also presented, and the

following options are allowed:

i) KO: PRINT SUMMARY: This key prints a summary of

the verification measurements on the current

hardcopy device,

ii) Kl: SAVE CONSTANTS: Pressing this key writes the

current n-test and calibration constants out to the

system status data file.

202
iii) K8: EXIT: Returns processing to the master

calibration supervisor.

10. If the newly measured n-test and calibration constants

are to be saved in a data file, press Kl: 'SAVE

CONSTANTS'. You will be instructed to insert the Binary

file loader disc in drive 1 and press 'CONTINUE' when

ready. The new constants will be written out to

SYSTATDAT, the system status data file, for use by the

n-test and calibration routines. Processing then

returns to the option menu. With a Winchester drive,

the new constants are written out to the system volume

and you will not swap flexible discs.

11. Pressing K8: 'EXIT' returns processing to the command

option menu of the master calibration supervisor.

203
INSTRUCTIONS FOR CREATING AND MODIFYING

STANDARD CALIBRATION DATA FILES

One of the considerable advantages of automated

calibration systems is their ability to perform

calibrations with instructions read from a data file on

disc. For Automated System II, this is available as an

option when an instrument calibration procedure is

initiated.

The routine to create and edit a standard calibration

data file is called "STDCALFILE" and is found on the

Utilities disc. To access this utility, press function

key K4 on setup menu II (Utilities) in the ac-dc program

supervisor. You will then be instructed to insert the

utilities disc in drive 0 and the target disc (one of

the instrument calibration master discs) in drive 1.

CONTINUE initiates the loading of the standard

calibration file program and its execution. If a

Winchester drive is designated as the mass storage

device, the data will be written out to a file on he

system volume of the drive and you will not swap discs.

Upon execution, STDCALFILE obtains a catalog of all

files on the target disc with the first two characters

S_; these denote standard calibration files. The catalog

of these files is displayed along with a function key

menu from which you may select a file operation. If

more than one calibration file is found, you will be

requested to type in the name of the standard

calibration file to be accessed. If no files are found,

the utility automatically assumes that a calibration

procedure should be created and branches down to do

this.

The top-level command option menu consists of the

following options:

i) KO: CREATE FILE: This option creates a standard

calibration procedure data file,

ii) Kl: MODIFY FILE: This allows the user to modify an

existing calibration procedure file,

iii) K2: CATALOG: Obtains a catalog of all the command

procedures on the target disc,

iv) K3: DELETE FILE: Allows deletion of standard

calibration files,

v) K4: NEW FILE: Deassigns the current data file and

allows another data file to be edited without

exiting the routine,

vi) K8: EXIT: Returns processing to the ac-dc master

supervisor.

vii) K9: ABORT MODE: Used to abort the present

processing mode and return control to the top-level

menu.

204
If a file is to be created (KO) , the system will ask you

to input a file name (up to eight characters) to be

appended after the "S_", then notify you that the file

will be created, and create the file on the target disc.

The system then displays the standard test parameter

input menu (the same one displayed during a

calibration) . The information that you supply will be

written out to the standard calibration file. This

process is repeated for all the voltage ranges to be

calibrated, and is aborted by the 'ABORT MODE' function

key (K9) . Processing is then returned to the top-level

option menu. During the data entry stage, the test

parameters may be edited by using the function key menu

presented after each voltage range is defined. This

menu operates in the following manner:

i) K1-K6: ACCEPT PARAMETERS: These keys accept the

information input to the screen menu,

ii) K3-K8: REJECT PARAMETERS: These keys reject the

current input test parameters and presents a second

function key menu which allows the operator to

modify any field in the parameter menu.

If the test parameters need to be modified, the

appropriate keys on the input modification function key

menu are:

i) K2: STANDARD ID: Allows modifications of the NBS

standard identification,

ii) K3: VOLTAGE: Allows the applied voltage to be

edited.

iii) K4: TEST RANGE: Allows the test unit voltage range

to be modified,

iv) K5: STANDARD RANGE: Allows editing of the standard

voltage range.

V) K6: FREQUENCY: Modifies the frequencies defined

for the standard calibration file,

vi) K9: ABORT MODE: Exits the command input mode and

returns processing to the standard calibration file

command menu. After you select the field to be

modified, the input cursor will appear at the first

character position of the field. You may then type

in the new data, overwriting the previous, erroneous

data. After the new data, are entered, the system

presents the updated test parameters for you to

accept or reject as before.

As each calibration point is accepted, the parameters

are stored internally in memory and this operation is

repeated until all the calibration points have been

defined. After the final calibration point has been

defined, press K9: 'ABORT MODE' to return processing to

the standard calibration routine command option menu.

205
8. After K9 : 'ABORT MODE' is pressed to exit the

calibration point definition routine, the program once

again presents the catalog of standard calibration files

and the option menu. The option keys are described

below.

9. If KO: 'CREATE FILE' is pressed, the program asks you

to supply the necessary file name and processing

continues as if no files were initially found. Again,

K9: 'ABORT MODE' is used to return to the command menu.

10. To modify a standard calibration data file, press Kl:

'MODIFY FILE'. The program will request the name of the

standard file to be modified and will present an

abbreviated listing of the file, showing the record

number of each record, the applied voltage defined in

that record, and the NBS standard TVC defined for that

voltage. A function key menu will be displayed across

the lower portion of the screen, with the following

definitions:

i) KO: VIEW RECORD: Permits the display of all the

information in a selected record,

ii) Kl: ADD RECORD: Permits the addition of a record

to the standard calibration data file,

iii) K2 : CATALOG: Displays a catalog of all currently

defined calibration data files on the disc,

iv) K3: MODIFY RECORD: Allows modification of a record

in the data file,

v) K4: DELETE RECORD: Purges a record from the

standard calibration file,

vi) K9: ABORT MODE: Returns processing to the

calibration file command menu.

lii Pressing any function key in this menu which accesses a

record results in an arrow (->) being displayed to the

left of the first record. To select the record of

interest, move the arrow to that record by pressing the

SHIFT key and rotating the knob clockwise to move the

arrow down and counterclockwise to move the arrow up.

When the arrow points to the record of interest, press

the ENTER key to continue the record operation in the

following manner.

12. When a record is being viewed, it is displayed in the

same format that was used to input the record in the

first place. After viewing the record, pressing K1-K6:

'ACCEPT PARAMETERS' returns processing to the record

modification menu.

13. To add a record, press Kl: 'ADD RECORD'. The usual

template appears and you may enter the pertinent

calibration data into the template in the normal manner.

After the parameters are entered, the data is written

out to file and processing is returned to the file

modification menu.

206
14. K2: 'CATALOG' obtains a listing of all standard

calibration files on the disc, but returns processing to

the master menu. To reaccess the file modification

facility, Kl: 'MODIFY FILE' must be pressed again.

15. To modify a record in the file, press K3 : 'MODIFY

RECORD'. The arrow is presented to the left of the

currently defined records, and the record to be modified

is chosen by using SHIFT-knob to place the arrow to the

left of that record and pressing ENTER. The contents of

the record are displayed in the usual fashion along with

the 'ACCEPT PARAMETERS -REJECT PARAMETERS' function key

menu. Pressing K1-K6: 'ACCEPT PARAMETERS' saves the

record with no modification. Pressing K3-K8: 'REJECT

PARAMETERS' results in a further function key menu with

the following options:

i) KO: TEST NUMBER: and Kl: INSTRUMENT ID: are used

when defining an ac-dc difference test and are not

used in this case. They are presented on this menu

because the same menu is used for calibration

definitions

.

ii) K2: STANDARD ID: This key allows you to specify

another NBS standard TVC.

iii) K3: VOLTAGE: Permits modification of the applied

voltage for this determination.

iv) K4: TEST RANGE: Permits you to modify the voltage

range of the test unit,

v) K5: STANDARD RANGE: Allows modification of the

standard voltage range definition.

vi) K6: FREQUENCY: Permits the addition and

modification of the frequencies defined for this

test. Pressing any of these keys displays the block

cursor at the first character position in the proper

field. After you enter the data and press ENTER,

the updated record will be displayed along with the

function key menu. If any more modifications are

required, press K3-K8: 'REJECT PARAMETERS' and

processing preceeds as just described. If the

record update is complete, press K1-K6: 'ACCEPT

PARAMETERS' to write out the updated information to

the file and return to the file modification menu.

16. 'DELETE RECORD' is used to purge a particular standard

test record from the file. After you press the function

key, the arrow is displayed to the left of the list of

records, and instructions on record deletion appear at

the bottom of the screen. To delete a record, use

SHIFT-knob to select the record to be deleted and press

ENTER. The record is deleted from the file and the data

file updated. THERE IS NO REDUNDANCY FOR THIS

OPERATION. NO SECOND CHANCES ARE GIVEN IF THE WRONG

RECORD IS SELECTED!

207
17. Should a standard calibration file need to be deleted,

press K3: 'DELETE FILE'. The program asks for the name

of the calibration file to be deleted, and purges the

file from the disc. AS WITH THE RECORD DELETION

FACILITY, THERE IS NO REDUNDANCY FOR THIS OPERATION.

The file is deleted and processing continues at the

standard calibration file option menu.

18. If a new file is to be edited, press K4 : 'NEW FILE'.

This key closes the I/O channel to the present data file

and returns processing to the catalog facility. If

calibration files have been previously defined, a

catalog of these files is displayed and the master menu

appears. If no standard files are available, then

processing continues as if the file should be created.

19. To exit the standard calibration file program, press K8:

'EXIT'. This returns processing to the ac-dc master

calibration program command menu.

208
INSTRUCTIONS FOR CREATING AND MODIFYING

STANDARD CORRECTION TABLE DATA FILES

An advantage of the new automated system is its ability

to store the corrections to the standard thermal voltage

and current converters in a file to be read by the

system upon beginning calibration. In the past, one had

to write a separate routine to accomplish the creation

of these files, since the Hewlett-Packard computers are

not set up to edit binary data files readily. The new

software, however, allows for interactive creation and

entry into such a data file without resorting to writing

a separate routine.

To access the standard correction file utility, press

K9 : 'MENU'. This brings up the second-level utilities

menu, which lists 'STAND. CORR' as K5. Press this key

to access the standard correction file utility. You

will be instructed to insert the utilities disc in drive

0 and the target disc in drive 1. After inserting these

discs, press CONTINUE to load the standard correction

file utility. If a Winchester drive is designated as

the mass storage device, the data will be written out to

the proper file on the system volume, and you will not

swap discs.

After the subroutine loads, a function key menu will be

displayed to allow you to select the operation desired.

The keys are defined as follows:

i) KO: CREATE MATRIX: Assists in the creation of a

new standard correction table,

ii) Kl: TE REPLACEMENT: Allows the updating of the

standard correction table should a thermal element

of one of the standard units require replacement,

iii) K2: MODIFY V RANGE: This key allows you to change

one or more ranges in the correction table; for

example, if a range resistor should change value,

iv) K3: STANDARD CURVE: Allows modification of the M

and N of one of the standards defined in the

correction table,

v) K4: PRINT MATRIX: Prints the correction matrix to

the screen or external hardcopy device,

vi) K5: GRAPH MATRIX: * Plots relationships such as

the standard characteristic curve or the

frequency-voltage response from the correction

table.

vii) K6: SYSTEM STATUS: Displays the present system

status

.

iii) K8: EXIT: Exits the standard correction routine.

Press the appropriate function key to choose the

desired operation. NOTE: the options denoted by * are not operational yet (12/08/1987)

.

209
4. Choosing KO: 'CREATE MATRIX' allows you to define a

standard correction file from scratch. After the key is

pressed, a heading is presented, and you are asked to

enter the name of the standard correction file to be

created. This file name can be up to eight characters

long, to allow for the 'C prefix which denotes a

correction file.

5. After entering the name, you will be prompted to enter

the number of rows (voltage ranges) and columns

(cardinal frequencies) defined in the table, whereupon

the program will again instruct you to insert the target

disc in drive 1 (unless a Winchester drive is

available), and press CONTINUE. The data file will be

created with the proper number of rows and columns.

6. Next, the program will prompt you for the frequencies to

be defined as cardinal frequencies. The prompts will

appear along the centerline of the screen together with

the frequency number currently being input. Enter the

frequencies in the blank provided beside the prompt.

The frequency may be up to six characters long in either

upper or lower case. NOTE, however, that each frequency

MUST be identified as either Hz, kHz, or MHz, so the

calibration routine can select the correct frequency

range. Since the program searches the frequencies

entered for the letters K and M (in either upper or

lower case) , these suffixes are both necessary and

sufficient to specify the frequency.

7. To permit information to be entered into the correction

table, an input menu is displayed on the CRT with the

heading STANDARD CORRECTION DEFINITION MENU. This menu

' is divided into the following fields:

i) Standard: The standard ID.

ii) M: The standard M value,

iii) N: The value for the standard N.

iv) Resistance (Ohms) : The resistance of the series

resistor in the TVC.

v) Voltage Range (V) : The voltage range of the

standard unit. The cardinal frequencies are listed

vertically at the center of the screen with the

blank fields beside them.

8. To enter the data, simply type in the information. You

will be limited to the number of blanks in the fields;

if this length is exceeded, only the n characters

represented by the blanks will be accepted. Pressing

ENTER after each field v/ill advance the cursor to the

next field. Continuing in this manner, the input menu

for each standard is defined in turn. Notice that, if a

standard thermal element appears more than once in a

correction table (for example, as part of a coaxial

set) , the M and N will be repeated in the proper fields

automatically, to avoid having to type these values in

again.

210
9. After the correction table is filled, a routine to check

the input data is run. This begins with a set of

instructions to follow for checking the correction

information. Pressing CONTINUE after reading these

instructions begins the checking routine. After

CONTINUE is pressed, the screen is cleared and the

currently defined correction table displayed. Because

the screen is too small to support the display of the M,

N, resistance, and voltage range for all the standards

at once, these quantities are displayed on a single line

at the bottom of the screen, along with the number of

the current record. In the table itself, each standard

unit is displayed vertically on the left-hand side with

its defined ac-dc corrections listed horizontally to the

right. The list of standards is followed by the line

'Exit to main menu' which allows you to return to the

main option menu.

10. The data for the individual standards are available for

editing by using the SHIFT-knob combination to move an

arrow vertically up and down the left- hand column of

the table. When the arrow is placed at a particular

record, the M and N values, along with the resistance

and voltage range of the standard unit, are displayed at

the bottom of the screen. Pressing ENTER will allow you

to edit the information in the record being indicated by

the arrow. Choosing the option 'Exit to main menu'

closes the correction file and returns processing to the

main option menu.

11. When the data for a standard are to be edited, the

original input menu is presented again for input. The

information must be retyped (except for the M and N) in

the manner above. The M and N values are changed by

another method described later.

12. The procedures described in steps 9-11 above are

repeated until the matrix is correct. Choosing the

'Exit to main menu' option will close the correction

file and return processing to the main option menu.

13. Occasionally, the ac-dc differences of one of the series

resistors of a coaxial TVC set will change in value or

the geometry of the resistors mounted in a multirange

instrument will change, affecting the ac-dc differences

of the unit. When this occurs, the correction table

must be updated to reflect these changes. This is

accomplished by pressing the K2 : 'MODIFY V RANGE'

function key in the standard correction master menu. If

no correction files have been defined on the target

disc, then an error message is displayed and processing

is returned to the master option menu. If only one file

is found, it is assumed to be the file to be modified

and processing proceeds at step 14 below. If more than

one correction file exists on the disc, then a catalog

of all the correction files on this disc is displayed,

along with an arrow at the left-hand margin of the

211
catalog. To choose the data file on which to operate,

use SHIFT-knob to move the arrow vertically to the file

to be modified and press ENTER. If none of the files

defined on the current target disc are of interest,

place the arrow beside the last entry in the table,

labeled 'No file. Return to command menu'. This aborts

the voltage range modification mode and returns

processing to the option menu.

14. After selecting the file to be modified, the routine

reads in the data from the disc and enters the

CORRECTION TABLE CHECK MENU routine. This works as

described in steps 9 and 10 above. After the desired

standard ID is chosen by using SHIFT-knob, the data

input menu is displayed. Since a change in the resistor

of a TVC will not affect the characteristics of the

thermal element itself, only the resistance (whose

change may be the reason for the ac-dc difference

change) and the new corrections need to be entered.

15. After the updated corrections for a particular range are

entered, processing returns to the CORRECTION TABLE

CHECK MENU, where another voltage range may be edited if

necessary, as will probably be the case with a coaxial

set. If another range needs to be modified, proceed in

the same manner as in step 14. If no more changes are

to be made, set the arrow to 'Exit to main menu'. The

updated information will be written out to the data

file, the file will be closed, and processing returned

to the master command menu.

16. If the characteristics of a thermal element in a

standard change, it will be necessary to update the M

and N for that unit. To do this in a previously defined

data file, press K3 : 'STANDARD CURVE' in the master

command menu. As before, a catalog of standard

correction files defined on the target disc will be

presented, and the proper data file is opened by using

SHIFT-knob to select the desired file.

17. After the data file to be updated is selected, the

program will request the ID of the standard thermal

element whose characteristics have changed. For a

coaxial set, one thermal element may be used with two or

more resistors, so the routine will search the

correction table for each occurrence of the thermal

element ID. For the NBS high frequency standards and

commercial multirange units, the characteristics will be

defined for one thermal element only. In any case,

enter the ID of the thermal element to be modified at

the prompt. You will then be prompted for its new M and

N values. After these values are entered, they are

written out to disc and processing returns to the main

command menu.

212
18. Should a hardcopy of the correction table be required,

press K4; 'PRINT MATRIX'. As usual, the program will

display the catalog of all defined correction data

files, and SHIFT-knob is used to pick the file to be

printed. Once the file is read in from disc, the

following menu is presented:

i) K1-K6: OUTPUT TO SCREEN: Directs the correction

table information to the CRT.

ii) K3-K8: OUTPUT TO PRINTER: Directs the correction

table information to the defined PRINTER IS device,

iii) K5: SYSTEM STATUS: Displays the present system

status

.

iv) K9: EXIT: Exits the 'PRINT MATRIX' routine and

returns processing to the main command menu.

19. If the output should be directed to the CRT, press Kl or

K6. The currently defined corrections will be displayed

on the CRT, along with the instruction to press CONTINUE

after viewing the matrix. CONTINUE returns the program

to the 'PRINT MATRIX' menu. To direct the output to the

current PRINTER IS device, press K3 or KB. This writes

the output to the printer and returns processing to the

'PRINT MATRIX' menu. Pressing K9 : 'EXIT' brings the

program back to the master command menu.

20. To return to the ac-dc program master menu, press KB:

'EXIT' in the correction file command menu. Processing

will return to the master command menu and you may

continue with calibrations, utilities, and other

options

.

21. When a thermal element is replaced (due to failure or

overloading) , the corrections for the standard units

using that particular thermal element will be altered,

particularly on the low voltage ranges where the

contribution from the resistor is relatively small. In

the past, a change of a thermal element meant a complete

redetermination of the corrections for that thermal

element. However, Hermach (19B4) has devised a

mathematical relationship linking the ac-dc differences

of the newly replaced thermal element with its previous

corrections. This process has been applied to the

software for the new automated system.

22. To enter the replacement routine, press Kl: 'TE

REPLACEMENT', in the main command option menu. This

will load and execute the TE replacement subroutine.

The subroutine begins by obtaining a catalog of all the

correction files on the target disc and prompting you to

select the file to be modified, as in the instructions

above. You will then be prompted to enter the ID of the

replaced thermal element. The program will scan the

data in memory to ensure that the present correction

table specifies this thermal element. If it does not,

you will be given the choice of accessing a new data

file or exiting the routine.

213
23. To update the correction table, the thermal element must

be checked both as a voltage converter and as a current

converter. The program will prompt you to enter the new

ac-dc differences as a voltage converter at the cardinal

• frequencies defined in the correction table. Next, you

will be asked for the voltage range at which the thermal

element was checked as a current converter and its ac-dc

differences measured in this configuration. Upon entry

of these data, the Hermach approach will be applied to

each voltage range of the standard thermal element and a

new correction table calculated.

24. The updated correction table will be written out to the

target disc and will also be available to be viewed and

printed as in steps 18 and 19 above. Processing then

returns to the main option menu.

214
INSTRUCTIONS FOR CALIBRATING AN ABBEY TE-2 0

1. This section covers the generic instructions for

calibrating an Abbey Electronics Model TE 2 0 thermal

transfer standard using "AC_DC_MSTR" . The same

calibration routine, with some minor variations, that is

used to calibrate a Ballantine 1605 is used to test this

instrument. It is assumed that the program is loaded

and initialized per the instructions in section I, and

that your last action was to press the 'BALL. 1605A'

option on the instrument calibration menu.

2. Following the instructions on the screen, insert the

flexible disc marked "BALLANTINE 1605A" in drive 0 and a

calibration data disc in drive 1. The Ballantine 1605

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the 1605 calibration

supervisory program. If a Winchester drive is

designated as the mass storage unit, then the data will

be written out to the proper volume on the drive and no

disc swapping is necessary.

3. The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N" , then processing continues at the 1605

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. To calibrate an Abbey TE-2 0, first remove the mercury

cell in the instrument case. To do this, remove the

screws holding the front panel to the case. The front

panel slides out of the casing, exposing the battery

holder and mercury cell. Remove the cell and replace

the front cover. The output EMF of the TE-2 0 is read

through a three pin Amphenol connector which, with pins

A and B used, is incompatable with the three pin adapter

made for the Ballantine 1398A coaxial units, and which

renders useless the normal method of using the analog

integrators. Instead, the output EMF is measured with a

nanovoltmeter , such as a Keithley 181. Use an Amphenol

three pin female connector to get onto the output

connector, and an appropriate cable to go to the

millivolt input connector of the nanovoltmeter. Make

sure the power switch of the TE-2 0 is on and that the

215
appropriate plugin resistor is employed. For an n-test,

the resistor should select the 100 volt range. Make

sure the resistor is in the correct orientation, usually

with the range marking reading up.

5. The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

these defaults. The options available are:

i) KO: STARTING VOLTS: The initial voltage for the

n-test (default is 70 volts)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 100 volts)

.

iii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage,

iv) K3 : VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

v) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8 : EXIT: Returns processing to the Ballantine

1605 supervisory program.

6. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

7. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions . The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification,

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

\ iii) The balancing process for the standard channel DAC.

Note that the 1605 output EMF is measured using an

external voltmeter, such as a Keithley 181, and no

balancing of the DAC is necessary. No activity will

be displayed in the DAC balance window; it it a vestige of other calibration routines.

216
iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation. When the voltage is brought up on the dc

calibrator, set the output switch on the front panel

to the 'ON' position. The system will wait for

things to settle, and during this time the output

voltage may be monitored with the dc voltmeter.

8. As the n-test routine runs, the values read from the

nanovoltmeter as well as the present test parameters,

are displayed in the appropriate places in the realtime

display. This feature is a nice advance over the older

software, in which test data tended to scroll upwards

and off the screen.

9. In the course of the n-test, the applied voltages are

set automatically, and the data collected and reduced

with no operator intervention. If a problem does occur,

however, there are two procedures to avoid (or at least

limit) any possible damage to the system and test units.

In the first case, the computer monitors the status byte

of the dc supply and its output voltage (via the dc

voltmeter) , before each reading. If the voltage exceeds

the nominal applied voltage, or if the dc calibrator

self-destructs, the computer will open the output relay

to disconnect the output voltage from the test unit.

For all system errors, an error message, with possible

remedial action and a prompt for instructions is

displayed across the bottom of the screen. Since the

display line is limited to 80 characters, the message

scrolls in from right to left. In most cases, you will

be given a choice of continuing the measurement process

or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9, labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9: (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

appropriate Winchester volume) , and the results will, if

desired, be printed to the current hardcopy device,

automatically

.

217
12. If a data file for this test number already exists, you

may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF. : Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status

.

iv) K3: HELP: Invokes the Ballantine 1605 help

facility.

v) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor,

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

the sequence, and return processing to the

Ballantine 1605 command option menu. If the SHIFT

key is not pressed, however, processing will be

halted upon completion of the test sequence, the

supplies will remain active, the output on, and

another option menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'BALLANTINE 1605' key, the system reads the command

procedure from the disc. If the 'BALLANTINE 1605' key

was pressed alone, however, an input menu, requesting

the standard unit, applied voltage, standard range, test

range, and desired frequencies is displayed, and the

parameters for the test are entered at this time. When

the output voltages are set on the ac and dc

calibrators, position the output switch to 'ON' and

check the output voltages on the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

218
i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE BALLANTINE

1605 COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the standard channel DAC can work without being

under- or over ranged. Press CONTINUE, when the switch

is set, to continue processing.

16. After processing is continued, the program will paint a

realtime display composed of the following five windows

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and do

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage.

17. If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

219
menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9: 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Ballantine 1605 command option

menu. Pressing SHIFT-K9: 'SOFT KILL' will halt

processing but leave the supplies and data acquistion

system unaffected.

18. During the calibration sequence, the data gathered from

the data acquisition system and nanovoltmeter will be

displayed in realtime. At the conclusion of each

; frequency, the ac-dc difference results for that

frequency will be sent to the currently active hardcopy

, device.

19. If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

20. When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the Ballantine

1605 command menu. If manual shutdown has been

specified, the following menu is displayed:

i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2 : INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4 : NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8: EXIT: Returns processing to the Ballantine

1605 command option menu.

21. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

220
i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined,

iii) The ac-dc corrections for the standard unit have not

been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

proper information and continue the measurement

process normally.

Upon exiting the n-test or calibration routine and

returning to the Ballantine 1605 command option menu,

the function keys defined in this menu may be used to

restart a test or to exit from the 1605 calibration

supervisor. If the latter action is taken, the

Ballantine 1605 subroutines are deleted from memory and

processing is returned to the ac-dc master program

option menu. From there, pressing any key will take the

action described by that key, and pressing 'EXIT' will

exit the program and return processing to the BASIC 5.0

system.

221
INSTRUCTIONS FOR CALIBRATING A BALLANTINE 1605A

1. This section covers the generic instructions for

calibrating a Ballantine 1605A automated multirange

thermal transfer standard using "AC_DC_MSTR" . It is

assumed that the program is loaded and initialized per

the instructions in section I, and that your last action

was to press the 'BALL. 1605A' option on the instrument

calibration menu.

2. Following the instructions on the screen, insert the

flexible disc marked "BALLANTINE 1605A" in drive 0 and a

calibration data disc in drive 1. The Ballantine 1605

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the 1605 calibration

supervisory program. If a Winchester drive has been

designated as the mass storage device, the data file

will be written to the appropriate volume, and this disc

swapping is unnecessary.

3. The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N", then processing continues at the 1605

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. To perform an n-test on a Ballantine 1605A, the

instrument must be set to the 64-125 volt range, and the

mode switch must be placed in the 'Manual Xfer'

position. The input voltage is applied through the 'Hi

Input' connector. Since the 1605 has a high output

impedance, the EMF cannot be measured using the analog

integrators in the normal manner. Instead, the output

EMF is measured with a nanovoltmeter , such as a Keithley

181. Use a double banana-to-twin Amphenol adapter to

connect the double banana output jacks to the millivolt

input connector of the nanovoltmeter.

5. The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

these defaults. The options available are:

222
i) KO: STARTING VOLTS: The initial voltage for the

n-test (default is 95 volts)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 125 volts)

.

iii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage,

iv) K3: VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

V) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8: EXIT: Returns processing to the Ballantine

1605 supervisory program.

6. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

7. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions . The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification.

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

iii) The balancing process for the standard channel DAC.

Note that the 1605 output EMF is measured using an

external voltmeter, such as a Keithley 181, and no

balancing of the DAC is necessary. No activity will

be displayed in the DAC balance window; it it a

vestige of other calibration routines.

iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation. When the voltage is brought up on the dc

calibrator, set the output switch on the front panel

to the 'ON' position. The system will wait for

things to settle, and during this time the output

voltage may be monitored with the dc voltmeter.

223
8. As the n-test routine runs, the values read from the

nanovoltmeter as well as the present test parameters,

are displayed in the appropriate places in the realtime

display. This feature is a nice advance over the older

software, in which test data tended to scroll upwards

and off the screen.

9. In the course of the n-test, the applied voltages are

set automatically, and the data collected and reduced

with no operator intervention. If a problem does occur,

however, there are two procedures to avoid (or at least

limit) any possible damage to the system and test units.

In the first case, the computer monitors the status byte

of the dc supply and its output voltage (via the dc

voltmeter) , before each reading. If the voltage exceeds

the nominal applied voltage, or if the dc calibrator

self-destructs, the computer will open the output relay

to disconnect the output voltage from the test unit.

For all system errors, an error message, with possible

remedial action and a prompt for instructions is

displayed across the bottom of the screen. Since the

display line is limited to 80 characters, the message

scrolls in from right to left. In most cases, you will

be given a choice of continuing the measurement process

or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9 , labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9: (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

proper Winchester volume), and the results will, if

desired, be printed to the current hardcopy device,

automatically.

12. If a data file for this test number already exists, you

' may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF.: Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status

.

224
iv) K3: HELP: Invokes the Ballantine 1605 help

facility.

V) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor,

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

the sequence, and return processing to the

Ballantine 1605 command option menu. If the SHIFT

key is not pressed, however, processing will be

halted upon completion of the test sequence, the

supplies will remain active, the output on, and

another option menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'BALLANTINE 1605' key, the system reads the command

procedure from the disc. If the 'BALLANTINE 1605' key

was pressed alone, however, an input menu, requesting

the standard unit, applied voltage, standard range, test

range, and desired frequencies is displayed, and the

parameters for the test are entered at this time. When

the output voltages are set on the ac and dc

calibrators, position the output switch to 'ON' and

check the output voltages on the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

225
iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE BALLANTINE

1605 COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the standard channel DAC can work without being

under- or over ranged. Press CONTINUE, when the switch

is set, to continue processing.

16. After processing is continued, the program will paint a

realtime display composed of the following five windows

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

V) Previous results for all frequencies at this

voltage.

17. If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

I status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9: 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Ballantine 1605 command option

menu. Pressing SHIFT-K9: 'SOFT KILL' will halt

processing but leave the supplies and data acquistion

system unaffected.

226
18. During the calibration sequence, the data gathered from

the data acquisition system and nanovoltmeter will be

displayed in realtime. At the conclusion of each

frequency, the ac-dc difference results for that

frequency will be sent to the currently active hardcopy

device.

19. If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

20. When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the Ballantine

1605 command menu. If manual shutdown has been

specified, the following menu is displayed:

i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4: NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8: EXIT: Returns processing to the Ballantine

1605 command option menu.

21. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined,

iii) The ac-dc corrections for the standard unit have not • been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

227
proper information and continue the measurement

process normally.

Upon exiting the n-test or calibration routine and

returning to the Ballantine 1605 command option menu,

the function keys defined in this menu may be used to

restart a test or to exit from the 1605 calibration

supervisor. If the latter action is taken, the

Ballantine 1605 subroutines are deleted from memory and

processing is returned to the ac-dc master program

option menu. From there, pressing any key will take the

action described by that key, and pressing 'EXIT' will

exit the program and return processing to the BASIC 5.0

system.

228
INSTRUCTIONS FOR CALIBRATING A FLUKE 54 OB

1. This section covers the generic instructions for

calibrating a Fluke 540B multirange thermal transfer

standard using "AC_DC_MSTR" . It is assumed that the

program is loaded and initialized per the instructions

in section I, and that your last action was to press the

'FLUKE 54 OB' option on the instrument calibration menu.

2. Following the instructions on the screen, insert the

flexible disc marked "FLUKE 54 OB" in drive 0 and a

calibration data disc in drive 1. The Fluke 54 OB

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the 540B calibration

supervisory program. If a Winchester drive is

designated as the mass storage device, then the

calibration program will load without user intervention.

3. The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n~test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N", then processing continues at the 540B

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

these defaults. The options available are:

i) KO: STARTING VOLTS: The initial voltage for the

n-test (default is 60 volts)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 100 volts)

.

iii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage,

iv) K3: VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

v) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

229
vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8: EXIT: Returns processing to the Fluke 540B

supervisory program.

5. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

Connect the standard channel detector cable to the

output of the TVC and set the detector switch to

position 3

.

6. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions . The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification,

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

iii) The balancing process for the standard channel DAC.

iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation.

7. When the voltage is brought up on the dc calibrator, set

the output switch on the front panel to the 'ON'

position. The system will wait for things to settle,

and during this time the output voltage may be monitored

with the dc voltmeter.

8. As the n-test routine runs, the values read from the

data acquisition system, as well as the present test

parameters, are displayed in the appropriate places in

the realtime display. This feature is a nice advance

over the older software, in which test data tended to

scroll upwards and off the screen.

9. In the course of the n-test, the applied voltages are

set automatically, the DACs balanced, and the data

collected and reduced with no operator intervention. If

a problem does occur, however, there are two procedures

to avoid (or at least limit) any possible damage to the

system and test units. In the first case, the computer

230
monitors the status byte of the dc supply and its output

voltage (via the dc voltmeter) , before each reading. If

the voltage exceeds the nominal applied voltage, or if

the dc calibrator self-destructs, the computer will open

the output relay to disconnect the output voltage from

the test unit. For all system errors, an error message,

with possible remedial action and a prompt for

instructions is displayed across the bottom of the

screen. Since the display line is limited to 80

characters, the message scrolls in from right to left.

In most cases, you will be given a choice of continuing

the measurement process or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9, labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9: (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

appropriate Winchester volume), and the results will, if

desired, be printed to the current hardcopy device,

automatical ly

.

12. If a data file for this test number already exists, you

may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF. : Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status

.

iv) K3: HELP: Invokes the Fluke 540B help facility,

v) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor,

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

the sequence, and return processing to the Fluke

540B command option menu. If the SHIFT key is not

231
pressed, however, processing will be halted upon

completion of the test sequence, the supplies will

remain active, the output on, and another option

menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'FLUKE 54 OB' key, the system reads the command procedure

from the disc. If the 'FLUKE 540B' key was pressed

alone, however, an input menu, requesting the standard

unit, applied voltage, standard range, test range, and

desired frequencies is displayed, and the parameters for

the test are entered at this time. When the output

voltages are set on the ac and dc calibrators, position

the output switch to 'ON' and check the output voltages

on the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE FLUKE 540B

COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

232
where the DACs can work without being under- or

overranged. Press CONTINUE, when the switch is set, to

continue processing.

16. After processing is continued, the program will paint a realtime display composed of the following five windows

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage

.

17. If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9: 'HARD KILLJ will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Fluke 54 OB command option menu.

Pressing SHIFT-K9: 'SOFT KILL' will halt processing but

leave the supplies and data acquistion system

unaffected.

18. During the calibration sequence, the data gathered from

the data acquisition system will be displayed in

realtime. At the conclusion of each frequency, the

ac-dc difference results for that frequency will be sent

to the currently active hardcopy device.

19. If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

20. When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

233
output relay, and return processing to the Fluke 54 OB

command option menu. If manual shutdown has been

specified, the following menu is displayed:

i) KO: REPEAT VOLTAGE*. This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4: NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8: EXIT: Returns processing to the Fluke 540B

command option menu.

21. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined,

iii) The ac-dc corrections for the standard unit have not

been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

proper information and continue the measurement

process normally.

22. Upon exiting the n-test or calibration routine and

returning to the Fluke 54 OB command option menu, the

function keys defined in this menu may be used to

restart a test or to exit from the 54 OB calibration

supervisor. If the latter action is taken, the Fluke

54 OB subroutines are deleted from memory and processing

is returned to the ac-dc master program option menu.

From there, pressing any key will take the action

described by that key, and pressing 'EXIT' will exit the

program and return processing to the BASIC 5.0 system.

234
INSTRUCTIONS FOR CALIBRATING A GUILDLINE 7100A

1. This section covers the generic instructions for

calibrating a Guildline 7100A automated multirange

thermal transfer standard using "AC_DC_MSTR" . It is

assumed that the program is loaded and initialized per

the instructions in section I, and that your last action

was to press the 'GUILD 7100A' option on the instrument

calibration menu.

2. Following the instructions on the screen, insert the

flexible disc marked "GUILDLINE 7100A" in drive 0 and a

calibration data disc in drive 1. The Guildline 7100

calibration disc contains the necessary routines to do a complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the 7100 calibration

supervisory program. If a Winchester drive has been

designated as the mass storage device, the data file

will be placed in the appropriate volume and the disc

swapping is unnecessary.

3. The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N" , then processing continues at the 7100

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. Since the Guildline 7100A does not require that the

thermal element characteristics be known, no n-test is

required. Pressing K2 : AC-DC DIFF will initiate a

calibration.

5. Initial setup of the Guildline 7100A proceeds as

follows. Connect a stable dc source to the dc input

terminals on the front panel, and the ac-dc output of

the comparator to the ac input terminals. Connect a Keithley 181 nanovoltmeter to the recorder output on the

rear panel. Follow the setup instructions in the

Guildline 7100A manual through section 3.4.2 (G) , with

the following exception. The dc toggle switch must be

in the "-" position if non-floating dc ballast supply is

used. For the calibration sequence, the meter range

switch should be set on the 1% setting for those

frequencies where the ac-dc difference of the unit is

less than 10000 ppm, and at 10% for frequencies above

235
about 50 kHz. Follow the instructions in the manual to

set up the high-voltage extender box if needed.

Nineteen subroutines will be loaded from the supervisory

disc, and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

•BALLANTINE 7100' key, the system reads the command

procedure from the disc. If the 'BALLANTINE 7100' key

was pressed alone, however, an input menu, requesting

the standard unit, applied voltage, standard range, test

range, and desired frequencies is displayed, and the

parameters for the test are entered at this time. When

the output voltages are set on the ac and dc

calibrators, position the output switch to 'ON' and

check the output voltages on the ac and dc voltmeters.

If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE BALLANTINE

7100 COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the standard channel DAC can work without being

under- or over ranged. Press CONTINUE, when the switch

is set, to continue processing.

236
After processing is continued, the program will paint a realtime display composed of the following five windov/s

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

ii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage.

If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9 : 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Guildline 7100 command option

menu. Pressing SHIFT-K9 : 'SOFT KILL' will halt

processing but leave the supplies and data acquistion

system unaffected.

During the calibration sequence, the data gathered from

the data acquisition system and nanovoltmeter will be

displayed in realtime. At the conclusion of each

frequency, the ac-dc difference results for that

frequency will be sent to the currently active hardcopy

device.

If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the Guildline

7100 command menu. If manual shutdown has been

specified, the following menu is displayed:

237
i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4 : NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8 : EXIT: Returns processing to the Guildline 7100

command option menu.

14. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The two most notable

occasions are:

i) The ac-dc corrections for the standard unit have not

been defined in a data file,

ii) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

proper information and continue the measurement

process normally.

15. After a set of determinations has been taken with the

CHANNEL REVERSE switch on the "NORM" setting, the 7100A

must be calibrated again, at the same points, with the

switch on the "REVERSE" setting. The ac-dc difference

of the unit at each frequency is the arithmatic mean of

normal and reverse points.

16. Upon exiting the calibration routine and returning to

the Guildline 7100 command option menu, the function

keys defined in this menu may be used to restart a test

or to exit from the 7100 calibration supervisor. If the

latter action is taken, the Guildline 7100 subroutines

are deleted from memory and processing is returned to

the ac-dc master program option menu. From there,

pressing any key will take the action described by that

key, and pressing 'EXIT' will exit the program and

return processing to the BASIC 5.0 system.

238
INSTRUCTIONS FOR CALIBRATING A HOLT MODEL 6A

1. This section covers the generic instructions for

calibrating a Holt Model 6A multirange thermal transfer

standard using "AC_DC_MSTR" . It is assumed that the

program is loaded and initialized per the instructions

in section I, and that your last action was to press the

'HOLT 6A' option on the instrument calibration menu.

2. Following the instructions on the screen, insert the

flexible disc marked "HOLT MODEL 6A" in drive 0 and a

calibration data disc in drive 1. The Holt Model 6A

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the Model 6A

calibration supervisory program. If a Winchester drive

is designated as the mass storage device, the data will

be written out to the appropriate volume, and you will

not need to swap discs.

3 . The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N" , then processing continues at the Holt 6A

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. To calibrate a Holt Model 6A the instrument must be set

to the 150 volt range, and the power switch must be

placed in the 'ON' position. The input voltage is

applied through the UHF input connector. Since the

Model 6A has a grounded output the EMF cannot be

measured using the analog integrators in the normal

manner. Instead, the output EMF is measured with a

nanovoltmeter , such as a Keithley 181. Use a

microphone-to-Amphenol adapter to connect the microphone

output connector through an appropriate cable to the

millivolt input connector of the nanovoltmeter. Also

note that the protection button will probably need to be

depressed throughout the calibration to prevent the

ac-dc relay from tripping the Model 6A into a standby

mode.

5. The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

239
these defaults. The options available are:

i) KO: STARTING VOLTS: The initial voltage for the

n-test (default is 120 volts)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 150 volts)

.

iii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage,

iv) K3 : VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

v) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8 : EXIT: Returns processing to the Holt Model 6A

supervisory program.

6. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

7. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions . The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification.

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

iii) The balancing process for the standard channel DAC.

Note that the Model 6A output EMF is measured using

an external voltmeter, such as a Keithley 181, and

no balancing of the DAC is necessary. No activity

will be displayed in the DAC balance window; it it a

vestige of other calibration routines.

iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation. When the voltage is brought up on the dc

calibrator, set the output switch on the front panel

to the 'ON' position. The system will wait for

things to settle, and during this time the output

voltage may be monitored with the dc voltmeter.

240
8. As the n-test routine runs, the values read from the

nanovoltmeter as well as the present test parameters,

are displayed in the appropriate places in the realtime

display. This feature is a nice advance over the older

software, in which test data tended to scroll upwards

and off the screen.

9. In the course of the n-test, the applied voltages are

set automatically, and the data collected and reduced

with no operator intervention. If a problem does occur,

however, there are two procedures to avoid (or at least

limit) any possible damage to the system and test units.

In the first case, the computer monitors the status byte

of the dc supply and its output voltage (via the dc

voltmeter) , before each reading. If the voltage exceeds

the nominal applied voltage, or if the dc calibrator

self-destructs, the computer will open the output relay

to disconnect the output voltage from the test unit.

For all system errors, an error message, with possible

remedial action and a prompt for instructions is

displayed across the bottom of the screen. Since the

display line is limited to 80 characters, the message

scrolls in from right to left. In most cases, you will

be given a choice of continuing the measurement process

or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9 , labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9: (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

proper Winchester volume), and the results will, if

desired, be printed to the current hardcopy device

automatically.

12. If a data file for this test number already exists, you

may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF. : Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status.

241
iv) K3: HELP: Invokes the Holt Model 6A help facility,

v) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor.

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

the sequence, and return processing to the Holt

Model 6A command option menu. If the SHIFT key is

not pressed, however, processing will be halted upon

completion of the test sequence, the supplies will

remain active, the output on, and another option

menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'HOLT MODEL 6A' key, the system reads the command

procedure from the disc. If the 'HOLT MODEL 6A' key was

pressed alone, however, an input menu, requesting the

standard unit, applied voltage, standard range, test

range, and desired frequencies is displayed, and the

parameters for the test are entered at this time. When

the output voltages are set on the ac and dc

calibrators, position the output switch to 'ON' and

check the output voltages on the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

242
iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE HOLT MODEL

6A COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the standard channel DAC can work without being

under- or over ranged. Press CONTINUE, when the switch

is set, to continue processing,

16. After processing is continued, the program will paint a

realtime display composed of the following five windows

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage

.

17. If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9 : 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Holt Model 6A command option

menu. Pressing SHIFT-K9: 'SOFT KILL' will halt

processing but leave the supplies and data acquistion

system unaffected.

243
18. During the calibration sequence, the data gathered from

the data acquisition system and nanovoltmeter will be

displayed in realtime. At the conclusion of each

frequency, the ac-dc difference results for that

frequency will be sent to the currently active hardcopy

device.

19. If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

20. When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the Holt Model 6A

command menu. If manual shutdown has been specified,

the following menu is displayed:

i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 320 characters

in length.

iv) K3 : FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4: NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) KB: EXIT: Returns processing to the Holt Model 6A

command option menu.

21. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined,

iii) The ac-dc corrections for the standard unit have not

been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

244
proper information and continue the measurement

process normally.

Upon exiting the n-test or calibration routine and

returning to the Holt Model 6A command option menu, the

function keys defined in this menu may be used to

restart a test or to exit from the Model 6A calibration

supervisor. If the latter action is taken, the Holt

Model 6A subroutines are deleted from memory and

processing is returned to the ac-dc master program

option menu. From there, pressing any key will take the

action described by that key, and pressing 'EXIT' will

exit the program and return processing to the BASIC 5.0

system.

245
INSTRUCTIONS FOR CALIBRATING A HOLT MODEL 11

1. This section covers the generic instructions for

calibrating a Holt model 11 coaxial thermal transfer

standard using "AC_DC_MSTR" . It is assumed that the

program is loaded and initialized per the instructions

in section I, and that your last action was to press the

'Holt Model 11' option on the instrument calibration

menu.

2. Following the instructions on the screen, insert the

flexible disc marked "Holt Model 11" in drive 0 and a

calibration data disc in drive 1. The Holt model 11

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the Model 11

calibration supervisory program. If a Winchester drive

is designated as the mass storage device, then the

calibration program will load without user intervention.

3. The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N", then processing continues at the Model 11

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

4. The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

these defaults. The options available are:

i) KO: STARTING VOLTS: The initial voltage for the

n-test (defaults are 70, 90, and 120 volts,

depending on the current rating of the thermal

element under test)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 100, 120, or 200 volts depending

on the TE)

.

iii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage,

iv) K3 : VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

246
V) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8: EXIT: Returns processing to the Holt Model 11

supervisory program.

5. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

Connect the standard channel detector cable to the

output of the TVC and set the detector switch to

position 3.

6. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions. The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification,

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

iii) The balancing process for the standard channel DAC.

iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation.

7. When the voltage is brought up on the dc calibrator, set

the output switch on the front panel to the 'ON'

position. The system will wait for things to settle,

and during this time the output voltage may be monitored

with the dc voltmeter.

8. As the n-test routine runs, the values read from the

data acquisition system, as well as the present test

parameters, are displayed in the appropriate places in

the realtime display. This feature is a nice advance

over the older software, in which test data tended to

scroll upwards and off the screen.

9. In the course of the n-test, the applied voltages are

set automatically, the DACs balanced, and the data

collected and reduced with no operator intervention. If

a problem does occur, however, there are two procedures

247
to avoid (or at least limit) any possible damage to the

system and test units. In the first case, the computer

monitors the status byte of the dc supply and its output

voltage (via the dc voltmeter) , before each reading. If

the voltage exceeds the nominal applied voltage, or if

the dc calibrator self-destructs, the computer will open

the output relay to disconnect the output voltage from

the test unit. For all system errors, an error message,

with possible remedial action and a prompt for

instructions is displayed across the bottom of the

screen. Since the display line is limited to 80

characters, the message scrolls in from right to left.

In most cases, you will be given a choice of continuing

the measurement process or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9, labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9; (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

appropriate Winchester volume) , and the results will, if

desired, be printed to the current hardcopy device,

automatically

.

12. If a data file for this test number already exists, you

may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF. : Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status.

iv) K3: HELP: Invokes the Holt Model 11 help facility,

v) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor,

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

248
the sequence, and return processing to the Holt

Model 11 command option menu. If the SHIFT key is

not pressed, however, processing will be halted upon

completion of the test sequence, the supplies will

remain active, the output on, and another option

menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'Holt Model 11' key, the system reads the command

procedure from the disc. If the 'Holt Model 11' key was

pressed alone, however, an input menu, requesting the

standard unit, applied voltage, standard range, test

range, and desired frequencies is displayed, and the

parameters for the test are entered at this time. When

the output voltages are set on the ac and dc

calibrators, position the output switch to 'ON' and

check the output voltages on the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6: CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

iv) K5: SYSTEM STATUS: Displays the present system

status.

v) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE Holt Model

11 COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

249
detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the DACs can work without being under- or

overranged. Press CONTINUE, when the switch is set, to

continue processing.

16. After processing is continued, the program will paint a

realtime display composed of the following five windows

(from the top)

;

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage.

17. If all goes normally, the system will test all

. frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

^ output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9: 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the Holt Model 11 command option

menu. Pressing SHIFT-K9: 'SOFT KILL' will halt

processing but leave the supplies and data acquistion

system unaffected.

18. During the calibration sequence, the data gathered from

the data acquisition system will be displayed in

realtime. At the conclusion of each frequency, the

ac-dc difference results for that frequency will be sent

to the currently active hardcopy device.

1:9. • If the system is running from a standard calibration

• V file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

continue at the next voltage range, with the automatic

warmup facility in effect.

250
20. When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the Holt Model 11

command option menu. If manual shutdown has been

specified, the following menu is displayed:

i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

iii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu,

v) K4: NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8: EXIT: Returns processing to the Holt Model 11

command option menu.

21. In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined,

iii) The ac-dc corrections for the standard unit have not

been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

proper information and continue the measurement

process normally.

22. Upon exiting the n-test or calibration routine and

returning to the Holt Model 11 command option menu, the

function keys defined in this menu may be used to

restart a test or to exit from the Model 11 calibration

supervisor. If the latter action is taken, the Holt

Model 11 subroutines are deleted from memory and

processing is returned to the ac-dc master program

option menu. From there, pressing any key will take the

action described by that key, and pressing 'EXIT' will

exit the program and return processing to the BASIC 5.0

system.

251
INSTRUCTIONS FOR CALIBRATING AN NBS TYPE COAXIAL UNIT

This section covers the generic instructions for

calibrating an NBS type coaxial thermal transfer

standard (such as Julie TNB-102s, Hermach-Engelhard

coaxial standards, etc.) using "AC_DC_MSTR" . It is

assumed that the program is loaded and initialized per

the instructions in section I, and that your last action

was to press the 'NBS TYPE' option on the instrument

calibration menu.

Following the instructions on the screen, insert the

flexible disc marked "NBS TYPE" in drive 0 and a

calibration data disc in drive 1. The NBS type

calibration disc contains the necessary routines to do a

complete calibration of this instrument. The data

generated by the program will be written out to the disc

in drive 1. Press CONTINUE to load the NBS type

clibration supervisory program. If a Winchester drive

is designated as the mass storage device, then the

calibration program will load without user intervention.

The program will ask for the test number of the present

test, and an identification of the unit under test. The

program will attempt to find a previously defined data

file (D_nnnnnnnn, where nnnnnnnn is the test number) for

this test. If a data file has not been created, or if

the one it finds is empty, the program will ask you if

an n-test should be performed. If you answer with with

a "Y", a data file is created, the test number and

identification are written out to the file, and the

n-test routine is loaded and executed. If you answer

with an "N", then processing continues at the NBS type

command menu, and you will be requested to supply the

proper test M and N during the calibration. If a data

file already exists, an option menu will be displayed.

This will be covered in more detail later.

The n-test routine will load 11 subroutines from disc

and set the default parameters. These are displayed on

the CRT along with a function key menu used to modify

these defaults. The options available are:

i) KO: STARTING VOLTS: The initial voltage for the

n-test (default is 70 volts or 170 volts, depending

on the current rating of the thermal element under

test)

.

ii) Kl: HIGHEST VOLTS: The final voltage for the

n-test (defaults to 100 volts or 200 volts,

depending on the TE)

.

ii) K2: VOLTAGE STEP: The step between V and dV. This

is defined as either a percentage of the currently

applied voltage, a percentage of the final voltage,

or a flat step. The default is .1% of the presently

applied voltage.

252
iv) K3: VOLTAGE INC.: The increment between successive

nominal levels of applied voltage (default is 5V)

.

V) K4: EXECUTE: Starts the n-test procedure,

vi) K5: PRINTER OFF: Disables printouts of each

individual voltage step (default is printer on)

.

vii) K6: HEADINGS OFF: Disables headings from being

printed on the hardcopy device (default is headings

on) . viii) K7: DEGREE OF FIT: Allows the characteristic

response equation to be fitted to any degree,

although a linear fit is generally good enough and

higher orders are not yet supported anyway,

ix) K8: EXIT: Returns processing to the NBS type

supervisory program.

5. If any changes to the default parameters are desired,

press the function key associated with that parameter.

A prompt will be displayed, and the value may be entered

in the blank space at the prompt line. The option menu

will then be redisplayed to request the next operation.

Connect the standard channel detector cable to the

output of the TVC and set the detector switch to

position 3.

6. When the 'GO' key is pressed, the IEEE-488 instruments

needed for an n-test (dc supply, dc voltmeter, and data

acquisition system) are initialized to the required

conditions . The screen will be erased, and a display,

divided into five "windows", will be painted to the CRT.

From the top, these windows display the following:

i) The test number, date, and test unit identification,

ii) The present voltage, output EMF, initial and final

nulled output EMF, and the incremental nulled EMF.

iii) The balancing process for the standard channel DAC.

iv) The applied voltage, output EMF, n, and the standard

deviation of the previous points in the n-test.

v) Finally, the fitted characteristic response

equation

.

7. When the voltage is brought up on the dc calibrator, set

the output switch on the front panel to the 'ON'

position. The system will wait for things to settle,

and during this time the output voltage may be monitored

with the dc voltmeter.

8. As the n-test routine runs, the values read from the

data acquisition system, as well as the present test

parameters, are displayed in the appropriate places in

the realtime display. This feature is a nice advance

over the older software, in which test data tended to

scroll upwards and off the screen.

253
9. In the course of the n-test, the applied voltages are

set automatically, the DACs balanced, and the data

collected and reduced with no operator intervention. If

a problem does occur, however, there are two procedures

to avoid (or at least limit) any possible damage to the

system and test units. In the first case, the computer

monitors the status byte of the dc supply and its output

voltage (via the dc voltmeter) , before each reading. If

the voltage exceeds the nominal applied voltage, or if

the dc calibrator self-destructs, the computer will open

the output relay to disconnect the output voltage from

the test unit. For all system errors, an error message,

with possible remedial action and a prompt for

instructions is displayed across the bottom of the

screen. Since the display line is limited to 80

characters, the message scrolls in from right to left.

In most cases, you will be given a choice of continuing

the measurement process or exiting the routine.

10. In the second case, for serious "total disaster" types

of faults, function key K9, labeled 'HARD KILL' will

immediately reset all supplies, open the output relay,

and return processing to the supervisory level option

menu. THIS KEY MAY BE USED AT ANY TIME IN THE

PROCESSING! For faults which are not so dramatic,

pressing SHIFT-K9: (K19) 'SOFT KILL' halts processing

but leaves the supplies and data acquisition system

unaffected. Processing continues at the n-test command

menu.

11. After the n-test is complete, processing will return to

the supervisory option menu for further choices by the

operator. In addition, the n-test data will be written

to the appropriate data file on disc (drive 1 or the

appropriate Winchester volume), and the results will, if

desired, be printed to the current hardcopy device,

automatically.

12. If a data file for this test number already exists, you

may run another n-test, or choose one of the other

options. These options include the following:

i) KO: N-TEST: This option initiates an n-test as

described above,

ii) Kl: AC-DC DIFF. : Initiates an ac-dc calibration of

the test unit.

iii) K2: SYSTEM STATUS: Displays the present system

status

.

iv) K3: HELP: Invokes the NBS type help facility,

v) K8: EXIT: Exits the routine and returns to the

ac-dc master program supervisor,

vi) K9: HARD KILL: THIS KEY WILL RESET THE VOLTAGE

SUPPLIES AND OPEN THE OUTPUT RELAY TO DISCONNECT THE

VOLTAGE FROM THE UNITS UNDER TEST,

vii) K19: SOFT KILL: Pressing the SHIFT key and K9

stops processing but leaves the supplies and data

acquisition system unaffected. If the shift key is

254
pressed simultaneously with the n-test or ac-dc

difference keys (SHIFT-KO or SHIFT-Kl) , then the

system will automatically reset the voltage

supplies, open the output relay upon completion of

the sequence, and return processing to the NBS type

command option menu. If the SHIFT key is not

pressed, however, processing will be halted upon

completion of the test sequence, the supplies will

remain active, the output on, and another option

menu will be displayed.

13. The next logical step is to run an ac-dc calibration,

and this is initiated by pressing Kl. Nineteen

subroutines will be loaded from the supervisory disc,

and several variable initializations and file

assignments made. If, in the original supervisory

option menu, the SHIFT key was pressed together with the

'NBS TYPE' key, the system reads the command procedure

from the disc. If the 'NBS TYPE' key was pressed alone,

however, an input menu, requesting the standard unit,

applied voltage, standard range, test range, and desired

frequencies is displayed, and the parameters for the

test are entered at this time. When the output voltages

are set on the ac and dc calibrators, position the

output switch to 'ON' and check the output voltages on

the ac and dc voltmeters.

14. If the calibration is to be made from a standard

calibration file, the system will now pause to let the

units under test warm up. The length of time allowed

for this warmup is range-dependent and varies from five

minutes at voltages of less than 60V to 45 minutes for

voltages of 600V and more. If the test is to be done

manually, a warmup menu is displayed to allow you to

select the proper warmup time as follows:

i) K1-K6; CONTINUE WITH MEASUREMENTS: Starts the

measurement sequence,

ii) K2-K7: START SEQUENCE: If no automatic warmup is

desired, this functions in the same manner as K1-K6.

If, however, the system is currently in the midst of

an automatic warmup period, this action aborts the

warmup and starts the sequence,

iii) K3-K8: AUTOMATIC WARMUP: When pressed, this action

starts the automatic warmup period for the present

voltage range.

iv) K5: SYSTEM STATUS: Displays the present system

status

.

V) K9: HARD KILL: ABORTS ALL PROCESSING, RESETS THE

SUPPLIES, AND RETURNS PROCESSING TO THE NBS TYPE

COMMAND OPTION MENU,

vi) K19: SOFT KILL: Stops processing but leaves the

supplies and data acquisition system unaffected.

255
15. In any case, processing resumes with the system reading

the EMFs from both the test and standard channel. If no

output is measured, the system presents an error message

and suggests corrective action. You will then be

instructed as to the position in which to set the

detector switch on the front panel. This action will

ensure that the output EMF is divided down to a level

where the DACs can work without being under- or

overranged. Press CONTINUE, when the switch is set, to

continue processing.

16. After processing is continued, the program will paint a

realtime display composed of the following five windows

(from the top)

i) The test number and instrument identification,

ii) Present test data (the ac-dc difference and dc

reversal data)

.

iii) A summary of the ac-dc differences for this set of

four determinations,

iv) A presentation of the test parameters (standard

unit, output EMFs, currently applied voltage and

frequency, etc.).

v) Previous results for all frequencies at this

voltage.

17. If all goes normally, the system will test all

frequencies at the present applied voltage without

operator intervention. The system will monitor the

status bytes of the ac and dc supplies, as well as their

output voltages (via the voltmeters) and will reset the

system if either output voltage exceeds the present

nominal voltage range. Any system faults, including

output voltages out of range, equipment faults, and so

forth, will be presented on the bottom of the display

and scrolled from right to left. These error messages

usually include several options (exit to the command

menu, continue with the sequence) as well as suggestions

for remedial action on your part. As always, function

key K9: 'HARD KILL' will disconnect the output voltage

from the units under test, reset the supplies, and

return processing to the NBS type command option menu.

Pressing SHIFT-K9: 'SOFT KILL' will halt processing but

leave the supplies and data acquistion system

unaffected.

18. During the calibration sequence, the data gathered from

the data acquisition system will be displayed in

realtime. At the conclusion of each frequency, the

ac-dc difference results for that frequency will be sent

to the currently active hardcopy device.

19. If the system is running from a standard calibration

file, at the completion of each voltage range you will

be prompted to reconfigure the test and standard units

for the next range. Upon pressing 'C, processing will

256
continue at the next voltage range, with the automatic

warmup facility in effect.

When the automatic calibration test sequence is

complete, or if the automatic shutdown option is in

effect, at the end of the final frequency on the present

range, the system will reset the supplies, open the

output relay, and return processing to the NBS type

command option menu. If manual shutdown has been

specified, the following menu is displayed:

i) KO: REPEAT VOLTAGE: This key repeats all

frequencies for the present applied voltage,

ii) Kl: REPEAT FILE: This key will force the system to

repeat the previous standard calibration file,

ii) K2: INSERT NOTE: Use this key to insert a note on

the hardcopy. The note may be up to 32 0 characters

in length.

iv) K3: FREQUENCY: This key allows the definition of

up to eight new frequencies to be calibrated at the

currently applied voltage. The frequencies are

input into the usual test parameter input menu.

V) K4: NEW VOLTAGE: This key presents the test

parameter input menu and allows the definition of a

new voltage range and frequencies. The supplies and

data acquisition system will be reset when this

option is chosen, and will be initialized to the new

test parameters.

vi) K8: EXIT: Returns processing to the NBS type

command option menu.

In addition to the prompts and options described above,

on occasion the system may need some information that

has not been previously defined. The four most notable

occasions are:

i) The standard M and N have not been defined in a data

file.

ii) No n-test was run on the test unit and therefore no

test M and N have been defined.

ii) The ac-dc corrections for the standard unit have not

been defined in a data file.

iv) A series voltage-dropping resistor is being used to

make the ac and dc supplies operate at a higher

voltage where they tend to be more stable. In any

of these cases, the system will prompt you for the

proper information and continue the measurement

process normally.

Upon exiting the n-test or calibration routine and

returning to the NBS type command option menu, the

function keys defined in this menu may be used to

restart a test or to exit from the NBS type calibration

supervisor. If the latter action is taken, the Holt

model 11 subroutines are deleted from memory and

processing is returned to the ac-dc master program

257
option menu. From there, pressing any key will take the

action described by that key, and pressing 'EXIT' will

exit the program and return processing to the BASIC 5.0

system.

258
GENERAL INSTRUCTIONS FOR PREPARING AC-DC TEST REPORTS

General Information

The ac-dc test report files are stored in the

subdirectory [.REPORTS] of the ac-dc directory on the

VAX. The reports are created and edited using EVE

(Extensible VAX Editor) , a text processing package

supplied by Digital Equipment Corp. A template for each

type of transfer standard (Fluke 540B, Holt Model 11,

etc.) that the Division has calibrated has been stored

in the reports subdirectory using the file nomenclature

' instrument. RPT' . The extension .RPT indicates a report

file and all new report files should bear this

extension. Report files are generally named for the

test number of the ac-dc test. In the case where no

test number has been assigned to a test, the file is

usually named to reflect the customer. The report file

should then be renamed to the test number after one is

assigned.

Generating New Reports

To log in to the reports subdirectory, find an

unoccupied terminal connected to the VAX. Turn it on

and press RETURN. The VAX will respond as follows (your

replies are in upper case)

Username: ACDC_REPORTS

Password: REPORTS

(note that the password is not echoed to the screen)

The message of the day will appear, followed

by a

$

The $ is the VMS prompt and indicates that the VAX is

ready to receive instructions. If you need to work with

an instrument template, type

$ COPY instrument . RPT filename. RPT

This copies the instrument template into a file which

you can now edit.

To enter EVE, type

$ EVE filename. RPT ( filename . RPT is the name

of the test report file)

If the report file has not been created, EVE will create

the file. If the file already exists, EVE opens the

file and displays the first page of the contents on the

259
screen. In either case the file is ready to be edited.

5. The EVE Users Manual is located in room A-144. The use

of EVE is explained quite well and these general

instructions will not go into all the gory details. A

few points, however, are worth mentioning:

i) Take note of what type of terminal you're logged

into. The VT240 in A-144 has a different keypad

than the Microterms (which are VTIOO clones) . The

manual provides instructions on how to use each type

of terminal.

ii) In addition to the documentation in the manual, help

may be obtained from EVE in two ways. On the VT240,

pressing the HELP key displays a representation of

the keyboard and a list of the defined control codes

and function keys. Pressing DO and then typing

'HELP' on the command line will list the editting

commands currently recognized by EVE. On the VTIOO

clones, HELP is assigned to key PF2 on the numerical

keypad, and DO is assigned to PF4

.

iii) There are several ways to end an EVE session. These

are

a) EXIT - This saves the text currently in the

buffer and exits EVE.

b) CTRL-Z - This is the same as EXIT.

c) QUIT - This exits EVE but does not save the text

currently in the buffer. If changes have been

made to the text in the buffer, you will be

given a chance to discontinue QUITting before

EVE throws out your text.

d) CTRL-Y - This crashes EVE, wipes out the buffer,

and returns processing to the VMS command level.

e) The system crashes - Who knows what happens?

6. If EVE has been exitted (EXIT or CTRL-Z) the text is

written out to a file which has a version number 1 more

than the version number of the file you loaded into EVE.

This means that if you are totally disgusted with what

you just did, the original file is not lost or written

over, and can be EVEed again.

7. QUITting flushes the buffer and leaves the file as it

was before the EVE session.

8. CTRL-Y and system crashes both should generate journal

files. These files contain a record of every keystroke

made during an EVE session. To reconstruct the file in

EVE after such an event, enter EVE by typing

$ EVE/RECOVER filename. RPT

You will be able to pick up more or less where you left

off (the last few commands may not make it into the

file) . Note, however, that in the event of a system

260
crash, the journal file may not make it to the disc. In

this case, you will have to start over at instruction 4.

To get a hardcopy of the report, type either

$ PRINT filename. RPT

or

$ LASER filename. RPT

In the first case, the output will be directed to the

Hewlett-Packard LaserJet printer located beside the

manual comparator. This printer may be set to either

cassette feed or manual single-sheet feed. See the

manual for instructions on how to use these modes. The

second command will send the output to the LaserJet

printers in the computer room. The first 60 lines of

your text will be printed on letterhead bond on the

printer on the right (on top of the CPU cabinet) , and

the remainder of the file will be printed on plain bond

on the printer on the left (atop the expander cabinet)

.

To send the hardcopy to the Printronix P-3 00 printer in

the computer room, type

$ PRINT/QUEUE=SYS$PRINT filename. RPT

To end the session on the VAX, you may log off using any

of these commands:

$ LO

$ LOGOUT

$ BYE

$ OFF

$ SCRAM

$ BUG_OFF

$ GET_LOST

The VAX will display the time of logoff and begin

ignoring you.

261
Section 15

INSTRUCTIONS FOR USING THE MANUAL AC -DC DIFFERENCE COMPARATOR SYSTEM

E. S. Williams and C. B. Childers

I . General

II. Power Supplies and Controls

III. Console

IV. TE Comparator

V. Galvanometer Circuits and Shields

VI. AC-DC Difference Tests

Contents

VII Grounding

VIII Calculation of nE

IX Low- Frequency Tests

262
I. General

This comparator system is used to make ac-dc difference tests of voltage and

current converters plus certain other tests. The nominal range of testing is

1 to 1000 V from 20 Hz to 100 kHz and 5 mA to 20 A from 20 Hz to 50 kHz;

however, with certain precautions and additional equipment, tests can be made

down to 2 Hz at low voltages and currents and up to 1 MHz from 1 to 100 V.

The standards are thermal voltage converters (TVC's) for voltage measurements

and single-range thermoelements (TE's) for current measurements. Their ac-dc

differences are determined by a series of special tests described in other

parts of this publication. The corrections are generally small and are known

over the full working range.

The TVC set (Model F, No. 7) consists of six series resistors and two TE's:

Fl (2.5 mA, 400 Q) and F2 (5 mA, 400 n) . Range combinations from 1 to 1000 V

are assembled as indicated by the labels on the eight units in the set.

For calibrations at currents up to 500 mA the TE's that serve as thermal

current converters (TCC's) are mounted on a four-pole rotary switch (in box

NBS 84). These TE's are all of the insulated-bead type, and all except the

500-mA unit are in vacuum enclosures. Separate unmounted TE's are used at 1 A

and higher currents

.

Calibration tests are occasionally made for instruments with a pointer-andscale readout. All of the above tests are usually made by observing and

evaluating small changes in thermal EMF's as ac and dc voltages or currents

are successively applied to the test and standard instruments connected

together. The equipment and procedures are described in the following

sections

.

263
II. Power Supplies and Controls

The voltage and current supplies, a matching output transformer, and remote

control connections are mounted in a rack which stands to the left of the

operator

.

Direct Voltage is supplied by a Cohu dc voltage calibrator, Model 324, which

has been modified by the addition of a remote control. The six control dials

are used when the local -remote switch is set at LOCAL. When the switch is

turned to REMOTE, the control shown in Fig. 1 is substituted for the three

lower control dials.

If another supply without fine controls is substituted for the regular one,

the series-resistance control shown in Fig. 4 may be inserted in series with

the output.

Direct Current is supplied by a Systron-Donner Model M5C15-50 TOV, with

voltage and current controls on the front panel. The front-panel controls for

current can be replaced by a 500-Q potentiometer connected across terminals 1 and 2 on the back of the supply (see Instruction Manual, Section 2.3). Fine

remote adjustment is provided by connecting 10 O across these terminals and

placing the circuit shown in Fig. 2 in parallel. One of four sensitivities

may be selected by means of a lever switch on the side of the control box.

AC Voltage and Current are supplied by an Optimation oscillator, amplifier,

and transformer system. Model AC- 104. The output of the oscillator (10 Hz to

100 kHz) is connected to the remote coarse and fine control shown in Fig. 3.

This circuit is used to adjust the input to the amplifier. The coarse control

is most satisfactory when it is operated near full range: 8 to 10 turns. The

oscillator gain control, therefore, should be set for an output level that is

only a little higher than adequate for the desired amplifier output.

Maximum rated voltage outputs for the seven transformer taps are listed in the

table below. The output is most stable at about 25% of these voltages; thus,

a tap about four times higher than the test voltage is recommended. The

amplifier does not perform well at the extreme frequencies. An additional tap

increase to about eight times the test voltage is suggested for tests below 20

Hz and above 70 kHz

.

The maximum current ratings for the seven taps are listed under "Current Mode"

below (Table 1). Current stability is satisfactory up to at least 50% of

maximum.

A small load should be kept on the transformer at all times when it is

operating in the current mode. Load resistors and switches are shown at E in

Fig. 6A. These are substituted for the current converter load by using the

rotary ac-dc switch at C in that figure. (See C and E in Section III.)

264
Table 1

Voltage Mode Current Mode Load Resistance (Cl)

16 V

40 V

80 V

160 V

400 V

800 V

1600 V

20

8

4

2 0.8

0.4

0.2

Short

During an intercomparison the ac and dc test voltages are adjusted to produce

nearly the same response (or output) of the test TVC by using the supply

controls. The final null adjustments can be made with the 10-turn, 1000-Q

helical potentiometer in the controller box. (See Fig. 4.) The 200-Q

resistor is connected in series to improve the linearity. The potentiometer

is shunted by one of ten selectable resistors of different values. The

minimum (nonzero) resistance is 10 fi (for low-voltage control) and the maximum

is about 820 n. A suitable control sensitivity can be found by selecting an appropriate shunting resistance.

Ten

^JM?_, Turns

JV-i- ^ 4- J I u I

Figure 1. DV Control

30 Q

-WW\A2 Q

200 Q

Three

Turns

Figure 2. DC Control

265
Figure 3. AV-AC Control

0

^wv' 510 Q

_yNWv" 270 Q

,—Awv. 150 Q

200 Q

820 Q

wwif 100 Q

62 Q

36 Q

20 Q

10 Q

1 kS2 Ten Turns

Figure 4. Series-Resistance Control

266
Ill

.

Console

Ac power is connected to the monitoring voltmeter and ammeter and the

galvanometer lamps as shown in Fig. 5. The ac-dc and av-dv test circuit

diagrams are shown in Figs. 6A and 6B.

The input to each circuit goes through a small bakelite panel on the left side

of the console. Ac voltage and current are connected through separate GR 874

coaxial connectors. Dc current is carried by a twisted pair of wires

extending from the power supply to the ac-dc switch and passing through a grommet in the input panel. Dc voltage is carried by a shielded pair of wires

connected to the terminals on the back of the power supply inside the rack.

These wires pass through a grommet in the input panel to two fuses mounted on

the panel.

The signals for ac and dc voltage and current are provided through coaxial

cables on the console front panel. The voltage cable (labeled "AV-DV")

terminates in a GR 874 connector, and a GR 874 tee is ordinarily used to

connect test and standard instruments together. The voltage circuit is

completely shielded if the test instrument has a coaxial input connector and

is itself shielded.

The current-carrying cable (labeled "AC-DC") has spade lugs for connecting to

current converter terminals

.

Certain features of these circuits are marked with the letters A through J on

the diagram and are explained in the following paragraphs.

A. This pair of terminals (labeled "DC") lets the operator insert instruments

or additional resistors in the current circuit. An ammeter or, if more exact measurements are needed (as in an n-test) , a shunt box may be

connected here. A resistor may be connected here also to better match the

total load to the 15 -V dc supply. Additional loading is provided on the

low-current ranges (1 A and lower) by a resistor added at the ammeter

input terminals. The additional load requires about a 2-V drop at rated

current and is used to improve the current stability.

B. This two-pin connector makes available to the operator the voltage drop across the ammeter plus the additional resistor. This potential could be

monitored during an n-test; however, it usually is not.

C. The ac-dc switch is a twelve-position, three-pole, enclosed silver-contact

switch with no stops. Ac and dc currents are switched to the test and

standard instruments in the normal test sequence: AC, DC+, DC-, AC. Two

sequences require one complete revolution of the switch. The switch is

usually turned in only one direction; however, it may be advantageous to use only one of the ac positions at high frequencies and high currents.

The latter procedure keeps the total impedance more nearly constant, and

less readjusting of the ac current is required.

267
D. The ac-dc switch has make-before-break contacts so that as the

instruments are disconnected, and while they remain disconnected, the

current supply voltage is applied at the points marked "D." If a different dc supply is used that requires a continuous load, as many

current -regulated supplies do, a shorting switch or a fixed load can be

added at these points.

E. The Optimation ac current supply (current mode) requires a very low

resistance load on the 20-A range and a small load on the other ranges.

Provisions have been made for a continuous load of 5 0, and this may be

shunted by 2 Q, 0.5 Q, or a short. Loads which have been found

satisfactory for each range are listed in Table 1, Section II. The

corresponding switches on the console front panel are labeled "AC load

(ohms)," and they are individually labeled "1.5," "0.5," and "0." These

are the approximate load resistances when that switch is closed. The ac contacts in the ac-dc switch connect these loads to the ac supply as the

instrument load is disconnected and while it stays that way.

F. After the test and standard TVC's are connected to the av-dv cable, av or dv is selected and dv is reversed by means of relays at F (Fig. 6B) . The

relays are powered by 24-V dc from a rectifier supply and activated by

microswitches which are closed by a cam on a stepping motor. The stepping

motor is advanced by closing a foot switch momentarily.

G. The av-dv circuit is activated by closing a four-position rotary switch,

which also turns on the 24-V rectifier as well as the left-hand amber lamp

to indicate that voltage is applied to the TVC's. This lamp should be on (with neither relay activated) during the preliminary adjustments, EMF

measurements, and warm-up. Switch G can also be used to insert a voltagedropping resistance of 900 or 1800 0 in series with the TVC's. This

additional resistance permits finer adjustment and promotes improved

stability for the lower TVC ranges (0.5 to 10 V) by allowing the ac and dc

power supplies to operate at an output of 10 V or more. The ac-dc

voltmeter (eleven ranges, 0.5 to 1000 V) indicates the nominal test

voltage at the av-dv cable.

Note: This voltmeter is not accurate when both the voltage and frequency

are high and may read low by as much as 50% at 50 kHz.

H. A red neon light on the front panel warns that hazardous voltage is

applied. It begins to glow at about 100 V and is brightly lit from 500 to

1000 V.

I. A plexiglass shield is placed between the operator's position and the test

and standard instruments to help prevent personal contact with the high

voltage. This is of particular value for instruments having binding-post

input terminals. A shock is also possible if the coaxial connectors are opened while voltage is applied. Removing the shield will create an open

circuit in the av-dv output lead at relay J. Hence, the switch at G

should be OFF, with all three indicator lamps out, before the shield is

removed.

J. This relay has been included in the above discussion.

268
So c ket

Vm

Am

Galvos.—

Galvo.

Ampls.

i3 I

120/6.3

Power

vy u lie [

( z)

OnOff

±

Figure 5. Console Power Switches

Am

178 Q

900 900 Q MM 1 M 1 AC-DC Vm

To DV

Supply ^ Coil

§for J

Fig. 6B AV DV Circuits

Figure 6. AC-DC and AV-DV Circuits

269
IV. TE Comparator

Various comparator circuits can be used to minimize the effect of power- supply

instability. In this comparator a voltage-divider circuit is used to balance

a fraction of the larger output EMF against the smaller one. The balance or null is detected by a photo-cell galvanometer. At the null the divider

setting corresponds to the ratio of the two EMF's. If the test EMF (E^-) is

held constant as ac and both polarities of dc are successively applied to the

TVC's or TE's, the standard EMF (Eg) will change as a function of the ac-dc

difference of the test instrument relative to the standard at the chosen

frequency. The resulting imbalance in the divider will then produce a change

in the detector deflection proportional to this ac-dc difference. Small

fluctuations in the power-supply output produce nearly equal proportional

changes in the two EMF's, and so the indication is not appreciably affected by

these fluctuations. The relative immunity to instability depends on how well

matched the time constants and response characteristics of the TE's are, but

the stabilizing effect of the divider circuit usually affords a significant

advantage. Also, this arrangement relaxes the requirement that the monitored

EMF be held so precisely constant.

In this comparator the higher of the two EMF's is connected across a 1000-fl

Kelvin-Varley divider (Ejj input) and the lower EMF (El) to the variable tap.

The potentiometer is connected to the test instrument output (E^-) , whether El

or Ejj, during an ac-dc difference determination. The test EMF is monitored

with the potentiometer by using the set key and galvanometer #1. Imbalance

EMF's in the divider circuit are measured with the #2 galvanometer by using

the read key.

The divider switch is moved from READ to CAL to introduce a 0.5-0 change in

the resistance of the divider circuit (AR/R = 500 ppm) , and the resulting

change in indication of galvanometer #2 is observed and used to compute the

scale sensitivity (K) in ppm/mm or percent/cm. The derivation of the

equations and an explanation of their practical applications are given below.*

The test procedure is outlined in Section VI.

A 2-0 AR can also be introduced. This resistance change and the n-test key

can be used in making determinations of n. (See the paper cited below in the

footnote for more information.)

*This derivation is from the paper presented as section 9 of this publication

(E. S. Williams, "Thermal voltage converters and comparator for very accurate

ac voltage measurements," J. Res. Nat. Bur. Stand., vol. 75C, pp. 145-154,

July-Dec. 1971). Figs. 6 and 8 of that paper show the circuit of the TE

comparator and complement the above discussion. Fig. 9 of the same paper is a photograph of the TE comparator.

270
Development of Equations"

The ac-dc difference of a TVC is defined as

8 = (i:

where Va is the a-c vohage and Va the average of the two directions of dc-vohage required to produce the same output emf. In the circuit of figure 6 the a-c and d-c voltages are adjusted to give the same emf

Et, of the TVC under test, as indicated by a null on

the detector with ki closed. The same voltages are applied to the standard TVC. Then,

(2)

where V'^i is the d-c voltage required to produce the same output emf of the standard as Va, and the subscripts, t and 5, refer to the test and standard instru- ments respectively. If V'^~Vd«l, then, closely

enough

From the definition of n given in the paper

Esa Esd

risEs.

With the polarities as shown in figure 6, N = mE,-E,

(3)

(4)

where N is the detector voltage with k2 closed and

ki open, and m is the divider ratio (with the detector

resistance Rd>> Ri).

Thus,

Sr=5,+

n-sE, (5)

where the subscript a and d have the same meaning

as before.

If a galvanometer is used as the detector, the current through it with k2 closed is, by Thevenin's Theorem,

gb- lE, — Eg

Re

(6)

where Re is the resistance of the galvanometer circuit with El and £, = 0. With Ei and held constant and

pRi inserted by means of Si,

lac niEJ{\ + p)-Es

r:

(7)

where R'g is the resistance of the galvanometer circuit.

Since mEi — E,<<E, and p<<l, we have to a

sufficient degree of approximation.

'gb

In Ejp

Re

(Db-Dc)

S

(8)

where and Dc are the resulting galvanometer deflections and S is the galvanometer current sensitivity. In the ac-dc test, with mE, constant,

Iga Igd — ' Da- Dd

Re

Thus, from (3), (8) and (9),

8t = 8,+

p(Dct-Da)

(9)

(10)

The test and standard TVCs may be interchanged

if Et < Es. If Et is appUed to the input of figure 8 and

held constant, with the detector and potentiometer con- nected as shown in the figure, a similar analysis leads

to the following equations:

8, = 8,+

nJEt

and

8r=8,-f

p(Dd-Da)

n,Dx

The characteristic n is determined from

^E|E

(11)

(12)

(13)

by applying known changes in input voltage and

observing the changes in output emf with a high resist- ance voltmeter, as described in the text. If a galvanometer is used as the detector instead of the voltmeter

its sensitivity is determined by inserting pR\ as described. From equations similar to (8) and (9)

^E, p^D

Es ~ D,

(14)

If the thermocouple resistance of the test TVC,

Rtc, is significant, p in equation (7) should be replaced

by p', the fraction of Ri + Rtc inserted by Si. Similarly

if the potentiometer resistance Rp is significant, p

in equation (14) should be replaced byp", the fraction of Ri + Rp inserted by Si.

(Paper 75C3&4-321)

* with a high impedance detector. With

a galvanometer the formula for 6^ is

too complicated to be useful.

* These equations were developed by F. L. Hermach.

271
These equations were derived on the assumption that only one detector would be

used. Equations (5) and (11) apply if the detector is a microvoltmeter , and

they differ only to the extent that either the test or standard instrument may

have the higher output (E^ or Eg)

.

The corresponding equations [Eqs. (10) and (12)], where one calibrated

galvanometer is the detector, are very similar except that EMF's do not appear

directly in them. Because we use two galvanometers, we can, and do, use an earlier and more familiar sign convention in the numerator: p(^si ~ ^d)/'^s^l'

Deflection changes in mm or cm (D^^ - D^j) are multiplied by a scale sensitivity

factor K to compute ac-dc differences in ppm or percent. It is more convenient to compute 2(0^ - D^j) or 2AD]^ and 2Di from four galvanometer

readings than to compute the actual deflection changes. Therefore, the value

of K is p/ns2D2, where p is the fractional change (500 ppm or 0.05 percent)

introduced in the galvanometer in the 2Di measurement and ng is the

characteristic n for the standard instrument.

rru V 500 500/n . .^u on • Thus, K = . ^ = ppm/mm with 2Di m mm.

2nD2^ 2T>i

^

If 2Di is 280 mm and n is 1.8 (as is typical), then K is approximately 1. If

a graph is made of 500/n vs E, one step in the calculation is saved.

Alternatively, K =

^2D^^^ percent/cm with 2D2^ in cm.

If 2Di is 28 cm, K is approximately 0.001.

272
V. Galvanometer Circuits and Shields

The comparator C, galvanometer amplifiers A, and secondary galvanometers G are shown in Fig. 7. The shielding and shield grounds are in dashed lines, and

switch Si is closed to connect the comparator circuit to the same ground.

Switch S2 and resistors 2 and 3 are used to reduce the read galvanometer

sensitivity.

Resistors 5 and 9 serve to insure a nominal external resistance for the

galvanometer amplifiers.

The other resistors provide proper damping and sensitivity for the secondary

galvanometers. They may be changed if different galvanometers are used. A

list of the present values of resistance is shown beside the diagram.

AD should be about 10 cm as the Lindeck potentiometer is turned on and off.

To check the

read galvanometer sensitivity

open

closed

down

2

"6 o'clock"

2 mV

30 cm

(use amp. zero control)

Read lever key medium (pull)

Test

(Top Light) Std. (Bottom Light) Ohms

1. 23

2. 750

3. 2150

4. 750

5. 274

! G G

L

6. 10000

7. 20000 2 3

8. 1000

9. 2370

I J

Figure 7

.

Galvanometer Circuits and Shields

273
VI. AC-DC Difference Tests

1. Connect the test and standard instruments to the power cables, and apply a relatively small dc voltage or current.

2. Connect the two output EMF's to the El and Ejj connectors on the TE

comparator, and bring up the dc input to the desired value while checking

the EMF's as in 4 below.

3. Interchange the connections if necessary so that the lower EMF is

connected to the El input and the higher one to Ey. Be sure the output

polarities are correct.

4. Measure both EMF's with the Lindeck potentiometer, using the set key and

the TGL (top galvo . light). See that both EMF's are in the normal range:

usually 7 to 10 mV.

5. Adjust the divider, using the read key, to set the BGL (bottom galvo.

light) near midscale. (If the EMF's are not connected as in 3 above, this

adjustment cannot be made. Occasionally the EMF's will be so nearly equal

that the divider cannot be** balanced with either connection. In this case

one EMF can be lowered by inserting a lOOO-O loading resistor between the

corresponding TVC or TCC output and the comparator cable.)

6. Adjust the gain on the right-hand photo-cell galvo. amplifier so that the

deflection change of the BGL is about 14 cm when the divider switch is

moved from READ to CAL with the read key closed. (If the EMF's are low,

the maximum deflection change may be less than 14 cm.)

7. Apply ac voltage or current at the test frequency, and adjust it to make

the output EMF's practically equal to those obtained with dc

.

8 . The test parameters may now be noted or measured and entered into the

computer. Apply dc for these measurements. After entering the voltage,

test range, and standard, measure Eg (EMF of the standard) as in 4 above

and enter its value. It may be El or Ej|. Measure E^ in the same way, but

do not enter it into the computer. Leave the potentiometer set at the E^

value for the remainder of the ac-dc difference determinations. Measure

l>l as in 6 above. The BGL may be moved down scale with the fine divider

control so that the deflections are to both sides of midscale. Record the

down-scale deflection (e.g., 18 cm), switch from READ to CAL (or vice

versa), and record the up-scale deflection (e.g., 32 cm). Switch back and

record another down-scale reading. The computer will then display the

test parameters and a computed value of K (in ppm/mm)

.

Reset the BGL to midscale and open the read key.

9. The computer will request the test frequency, which is entered in kHz.

10. Two ac-dc difference determinations can now be made by entering seven galvanometer readings, as requested on the monitor, in the sequence AC,

DC, DC, AC, DC, DC, AC as follows. (These readings should be made at

equal time intervals .

)

274
A. Apply ac voltage or current, and adjust the ac controls for a null on the

TGL using the set key. (The null position may change, so it should be

checked before most settings.) Hold both keys down (the read key may be

locked), and read the BGL deflection while keeping the TGL at null. Enter

the three-digit reading at the terminal, and unlock the read key.

B. Switch to dc and adjust the dc controls for a TGL null using the set key.

Read the BGL with both keys closed. Enter the reading at the terminal,

and unlock the read key.

C. Switch to reversed dc and repeat B.

D. Switch to ac and repeat A.

E. Repeat B, C, and D to complete the two determinations. The computer will

then display the data and the computed values of ac-dc difference.

275
VII. Grounding

The test and standard instruments are grounded at the ac source. The

switching circuit has a direct connection to ground with no switch contacts in

that line. If a switch were used in the ground line, and if it opened an

instant before the one in the high side opened, the TE bead could be damaged

by excessive voltage.

The two cables that connect the TVC's to the comparator have insulation

between the connector shell at the TVC end and the cable shield. This

insulation helps to prevent the TE heater current from returning to ground by

any route except through the TVC casing.

The comparator and galvanometer casings are grounded directly to the

laboratory power-mains ground. Their circuits are connected to the detector

case through the switch next to the detector connector. If the test TVC has

one side of its EMF output grounded, as a few of them do, the grounding switch

should be opened; the comparator and detector circuits are thereby grounded at

the test TVC.

The existence of an internal ground on the output of the test TVC is indicated

if an ohmmeter shows continuity between the EMF output terminal and the

instriiment case or low input terminal. The resistance would be very low, of

course, since the thermocouple itself is connected to the low input terminal

in this situation.

Note: It is advisable to use a high range on the ohmmeter to minimize the

current in the thermocouple circuit during this continuity check.

Current converters with binding posts for the input may be affected by the

magnetic field around the terminals. Additional voltage is often induced in

the thermocouple circuit, causing extra heating, when high currents are applied (5 A and higher). A coaxial lead about 10 cm long, therefore, should

be used between the test and standard instruments at these currents. Any loop

at the terminals should be minimized. When a shunted TVC is used as a standard, the shunt casing should be connected to the high or ungrounded line

from the power supply.

The figure below illustrates the proper connections.

Supply

Std Shunt n

TVC

Figure 8. Shunt Connections
VIII. Calculation of nE

n = M-(E*N)

A straight line (A-B) is drawn through the most-used portion of the n vs E

plot and extended to the intercept M at E = 0. The slope N is found by

reading the difference between M and a point on the A-B line at one of the

higher values of E and then dividing the difference by that value of E. In

the illustration below, M = 2.06 and the difference is 0.29 at E = 9. Thus, N = 0.032.

The product nE appears in Eqs. (3), (5), and (11) that were derived earlier in

Section IV. To illxistrate the use of M and N, a sample calculation of K = nE

for E = 7 is shown here.

k' = E*(M-E*N) = 7*(2. 06-7*0. 032) = 12.85 = nE

Figure 9. Plot of n vs E

277
IX. Low- Frequency Tests

Our calibration service extends down to 2 Hz at low voltages and low currents

(not more than 50 V and 50 mA) . Most of the standard TE's have significant

ac-dc differences at very low frequencies. The model C (MHz) TVC's with UHF

thermoelements (10, 20, 50, and 100 V) have large corrections because their

very short heaters permit substantial cooling between the peaks of ac current.

The standard TE's are much better, but still require corrections at 2 Hz.

Corrections for the low-mA standards vary with current rating and should

increase at higher currents where the heaters are shorter. All TE's perform

better at lower- than- rated current because the effect of cooling between ac peaks decreases rapidly as the heater current is lowered.

A special converter assembly has been built for low frequencies which takes

advantage of the better performance at low currents. It is designated "FE"

and consists of four 5-mA TE's (125 each) in series. It is ordinarily

operated at 2.5 mA, so that each element has an output of about 2.5 mV. The

thermocouple outputs are connected in series to give an overall output of

about 10 mV.

Since the assembly has an input resistance of 500 0, it is rated at 1.25 V at

2.5 mA. The table below lists rated voltages for the low-range TVC resistor

sets no. 1 and no. 7.

Set No. Rating (V) Resistance (0) Volts with FE

12

3

5 10

20

30

50

100

300

500

900

1900

3900

6 k

1.5

2.0

2.5

3.5

6

100

3-6

10-20

30-60

100-200

12 k

40 k

10 k

20 k

800

3600

11

16

26

51

3

10

31

101

278
Section 16

INSTRUCTIONS FOR USING THE MJTC COMPARISON EQUIPMENT

F. L. Hermach - April 1981

(revised Nov. 1982, July 1985, April 1986, and Nov. 1986)

I. GENERAL

This equipment is set up in Room A144 Metrology Building to compare the ac-dc

differences of multifunction and single- junction thermal converters (MJTC

s

and SJTC's). The standards and comparator are described in NBSIR 84-2903, the

1966 and 1976 papers included with it, and the 1986 CPEM paper [l]-[4]. You

should read these papers and sections II through VII of the IR before using

the equipment. The following instructions give additional information on the use of the comparator and on techniques for making comparisons. (The logbooks of Don Flach and the data books of F. L. Hermach, on which the 1976

paper on the MJTC's was based, are also in A144, along with the more recent

data, math, and summary sheets on which the IR was based.)

The MJTC's are as described in the 1976 paper, covering the range from 5 to 12

V at 30 to 10,000 Hz with 0.5-ppm accuracy. Their series resistors (for lowfrequency compensation) can be connected in parallel with the heaters to give

10- to 100-mA standards. The MJTC's are our primary ac-dc standards, so take

extreme care to avoid burning them out.

Two 5-mA thermoelements (TE's), labeled FX2 and FY2 , were evaluated with the

MJTC's to 10 kHz, as were the 5- and 10-V thermal voltage converters (TVC's)

formed from these TE's and the 5- and 10-V series resistors of the Fi set

described in the 1966 paper. The ranges have been extended in frequency,

current, and voltage by an involved process, which is described in the IR.

These TE's (with the F^ set of resistors) and a set of Weston TE's of very

special construction (to our specifications) from 10 mA to 20 A are our reference standards. They should be used primarily to test our working

standards. Their assigned ac-dc differences (5's) and other pertinent data

are filed in the ac-dc reference standards notebook in A144.

Section III and figures 5 and 6 of the IR describe the basic comparison

circuit. Figure A of these instructions shows the detailed wiring of the

equipment, including the important shielding and grounding. Figure B shows

the actual wiring of the EMF comparator; C the simplified schematics; D the

ac-dc relays; and E the input connections at the tank to guard against

burnout. Appendix 1 is a log of significant changes in the apparatus, which

should be continued as necessary.

II. AC -DC STANDARDS

III. COMPARISON EQUIPMENT

279
Stable ac and dc supplies are vital to high precision. The Model 303B Kintel

dc voltage standard, RCD-9 Optimation oscillator, and PA 50 T Optimation

amplifier are the same ones used in 1973-1976. The output transformer in the

PA 50 T is useful only to 30 kHz and has been replaced by an external KrohnHite Model MT-56R matching transformer. (Note the information on the

transformer panel.) The circuits of the simple fine-control units for the ac and dc supplies are shovm on their boxes. The ac-dc relays (which have an off

time of less than 0.5 ms) have a safe voltage limit (with resistive loads) of

about 100 V and a safe current limit of 50 mA (not simultaneously) . Spare

relays are available for experiments. You could use higher voltages with

another fast- acting switch, but you would have to take proper precautions

against electric shock.

A Keithley Model 150B chopper-type dc microvoltmeter is normally used as the

detector (N in figures B and C) , with the filter switch IN and the input

resistance set at 100 K. It is satisfactory if its amplifier is not

overloaded by ac from the thermocouple outputs (which can be especially

troublesome with MJTC's). Precautions and checks are given in appendix 2,

The output of the detector is connected to an electronic "averager" (op-amp

integrator and timer) , which integrates the output for 10 s to reduce the

effect of noise and to ease reading. Information on it is given in the ac-dc

data book, vol. 1, Nov. 30, 1981.

IV. n- COMPENSATION AND THE BASIC EQUATION

The EMF comparator balances a fraction of the EMF of one thermal converter

(TC) against the smaller EMF of the other. When both TC's have the same value

of n (e.g., two MJTC's), the balance is independent of small changes in the

supply voltage, which greatly increases the precision of the comparison. If,

however, the two n's are different, some of this advantage is lost. Properly

adjusting the added resistor Rj- in figures B and C restores this compensation,

as described in section III-A and appendix 1 of the IR.

The basic equation for this comparator is L = - 8^ = (N^j - Na)/ngE]^, where

t and s refer to "test" and "standard," respectively, and fi^- is the ac-dc

difference of the TC to which the Lindeck potentiometer is connected.* N^ is

the average of the two detector readings with ac applied in a sequence of four

ac and dc measurements, and N^ is the average with the two polarities of dc

applied in succession. Ei is the lower thermal EMF. If N is in nanovolts and

El in millivolts, A is in parts per million.

The "selection rules" are: 1) connect the TC with the higher EMF to E^ on the

comparator, and 2) connect the potentiometer to the TC with the higher n

(either one if their n's are equal). The "test TC" (6^-) is always the one to

which the potentiometer is connected, and hence is the one with the higher n.

As shown in appendix 1 of the IR, this equation applies when there is no current through the detector. For greater stability the 150B is generally

operated with a 100-kfl input resistance. This causes a negligible error when

two SJTC's are compared in the Read- 2 position of the divider, but the error can be 4% of A = 5^ - when two MJTC's are compared in the Read-1 position.

However, A is generally only a few ppm in such tests. If it is larger than 10

ppm, switch the 150B input resistance to the OPEN position.

280
V. OPERATION OF THE EQUIPMENT

A. Connections

For TVC comparisons connect the inputs of the TVC's in parallel by means of a GR 874-T connector. Lead impedances can cause serious errors at high

frequencies, so the center of the tee is arbitrarily but reasonably chosen as the reference plane for a comparison. (See the math notebook, Aug. 22, 1982,

for information on an unusual error caused by skin effect in the large

diameter center conductor of the GR 874 connectors.) For low-voltage ranges

put a higher-range Fi resistor in series with the pair of TVC's so that the

supply voltage will be about 5 V, for greater stability of the ac and dc sources. Plug one of the Zener-diode pairs, with a voltage knee at about

twice the input voltage V^^, across the input at the tank, with an expendable

TE in series, to protect against burning out the TE's of the TVC's. See

figure E,

The KH matching transformer should be used with the ac amplifier to avoid

large (and slow) dc transients and errors from the dc offset voltage Vq of

many direct-coupled amplifiers. If the transformer is not used, make sure that Vq is less than 0.1% of Vj^^. (See appendix 2 of the IR re this error.)

Note that the KH transformer has a high and a low frequency range. It is good

practice to connect an oscilloscope (with floating input) across the Zenerdiode (input) posts at the tank. Use it to check the frequency (roughly) with

its calibrated time scale and to look at the waveform, noise, and pickup.

Connect the outputs to the E^ (high EMF) and Ej^ (low EMF) jacks of the

comparator, and check all the other connections.

For comparison of two TE's as current converters (note the 50-mA limit)

connect their heaters in series. Shunt capacitance from either line (one line

is grounded) to the center conductor can cause errors at high frequencies.

This is best checked by repeating the test with a different lead arrangement.

Bead conductance (see section V of the IR) can cause errors that are best

checked by repeating the test at the highest frequency with the input leads

interchanged

.

B. Preliminary Adjustments

Make sure that the ac and dc switches on the relay box are off and that the

range switches, etc. on the supplies are properly set. Turn on the relay

power switch at the back of the box. (Note that if the power switch is off, ac is always applied when the ac switch is on.) Remove any shunting resistor

across E]^. Turn on the power supplies, detector (ac operation), averager,

oscilloscope, etc. Set the ac-dc switch to DC, turn on the dc switch, and

bring up the voltage to the desired value, checking the EMF's of the TC ' s on

the detector with the set key down. Check l^f. with the Weston Model 1 dc

milliammeter as you bring up the voltage. Make sure it is on the right range.

Make sure that the detector deflects upscale for each TC (or the signs will be

wrong). Reverse polarities at the output jacks on the tank, if necessary.

Then measure the EMF's of the TC's, with the potentiometer switch at E]^ and

then El and the divider switch at Pot'r. (Otherwise the divider loads the TC

output.) Record the EMF's and other data. (See the sample data sheet.)

281
Next switch the divider to Read-1 for two MJTC's (or an MJTC vs an SJTC) and

to Read- 2 for two SJTC's, and switch the potentiometer to the TC with the

higher n. Set the first divider dial to 0 when in the Read-2 position. Turn

on the potentiometer to an appropriate range and step down to the l-/iV range

of the detector, adjusting the potentiometer and the divider as needed, with

the set and read keys down, respectively. Because the divider loads E^, it

may be impossible to get a read balance when the two EMF's are nearly equal.

In that case connect a resistor whose resistance is approximately equal to the

input resistance of the divider (1 kfi in the Read-2 position and 10 kO in

Read-1) across the Ej^ terminals. Measure the new voltage Vj^ and use its value

in place of Ej^ in the basic equation, but use the value of n corresponding to

El if this is the standard TC. (See the math notebook. Mar. 17, 1982.)

C . n-Measurement

If it is not already known, measure ng (after warm-up) by changing the dc

input voltage by AV and observing on the detector the change AEg in the set

position (with the divider switch at Pot'r). Compute ng = VAEg/EgAV. (Set

AV/V to about 5 x 10"^.) To avoid errors from drifts, return the dc voltage

to the original value, check Eg after the same time interval, and use the

average value as the reference for AEg . Set the same AV/V in the negative

direction, and take the average of the two values of n. Unless the heater of

the TE is of Evanohm or its equivalent, there may be special problems in

defining and measuring n to 1%, because of the temperature coefficient of

heater resistance. (See appendix 1-D of the IR.)

D. n-Compensation (not needed for two MJTC's)

With dc applied, adjust the potentiometer until N < 0.1 /xV (set key down) and

the divider until N < 0.1 /zV (read key down). Then increase the dc voltage by

about 0.1% and adjust Rj- to give the same reading to within 5 ppm of Eg with

the read key down as before. Return the dc voltage to the original setting

and check the read balance. If balance is not achieved with Rj- = «> (disconnected) , then the other TC probably has the larger n. Switch the

potentiometer to it (it is now the test TC) and rebalance the potentiometer

and divider. Then proceed as before.

E. DC Reversal Compensation

Reverse the dc voltage and note the change in the detector reading with the

read key down. If it is more than about 20 divisions on the upper scale (1/5

full scale from side to side) on the desired detector range, switch on the dc

reversal compensator (RC) and adjust it to cut the change in half. The

compensator is wired (see the diagram on the box) so that it may be operated

in the same sequence as the main ac-dc switch. If the polarity is wrong

(opposite sequence) , simply rotate the switch knob 180° to reverse the

voltage. Reset the detector to midscale by adjusting the divider. Adjust the

potentiometer (set key down) so that nearly equal and opposite deflections are obtained for the two polarities of dc voltage.

282
F. AC Setting

Set the ac frequency, turn on the ac switch on the relay box, and switch to

AC, with the RC set to AC (i.e., off). Bring up the ac voltage while

monitoring the detector (set key down) , adjust the voltage to make N = 0

(midscale) , and check the read balance.

G. AC-DC Determinations

Always double check to make sure that the frequency setting is correct, that

the correct TC is recorded as the test TC on the data sheet, and that the

correct range is set on the detector.

For a determination, switch to AC, DC, DCR, and AC. In each case, (a) first

check the balance with the set key down and reset as necessary by adjusting

the voltage (center the ac and split the dc to within 40 ppm of E^- to keep the

An error less than 0.1 ppm)*; (b) then about 30 s after switching, hold the

read key down and reset the averager to 0.0; (c) at 40 s after switching,

start the averager; and (d) when its light goes out, release the read key,

rotate the ac-dc and RC switches, and record the reading of the averager. Do

not adjust the potentiometer or divider controls.

Repeat this sequence (or its inverse) three more times, with 30 s between the

start of the last AC in one sequence and the start of the first AC in the next

(so that a correction for nonlinear drift can be applied later if desired)

.

If the sequence timing is off, abort the sequence. If the interval timing

(between sequences) is off, record it and proceed,

H. Recording and Computing Data

See the sample data sheet for a convenient data- logging format. For each

sequence, 5^ - = K(-Dacl + D^c + ^dcr ~ ^ac2) • factor K = (F/2) /

(ngE]^), with El in mV. F/2 for the Model 150B detector is 10 times the range

in /iV (i.e., 10 for the l-/iV range). D is the reading of the averager (with

its sign) in divisions.

I . Shutting Down

Turn off the ac and dc switches on the relay control box, reduce the ac and dc

voltages, then turn off the ac and dc power supplies, oscilloscope, detector,

averager, potentiometer, and relay power supply (at the back of the box), and

turn the reversal compensator to AC (battery off)

.

If Rj. is adjusted so that AN in section V-D is less than 5 ppm of Eg for a 0.1% change in V^^^, then by direct proportion the error will be less than 0.1

ppm for a 20-ppm missetting of ^'^ test. This is 40 ppm of E^ for a nominally square-law TC (i.e., 0.4 if E^- = 10 mV) .

283
VI. MAINTENANCE OF THE SETUP

A. Inspections

1. Check the detector battery before each use. (The 150B will not

operate properly even in the ac mode if the battery is way below par.)

2. Check the various connections, especially the ground lead to the

conduit under the table, if erratic readings, sensitivity of the

detector to body position, etc. occur.

3. If any doubt about the RC (reversal compensator) arises, set 1000 div

on it and measure its output on the detector with the read key down.

It should be about 13 /zV with the Ei input shorted.

B. Periodic Tests (with the 150B detector)

These tests (recommended every six months) can be made by comparing items of

the measuring equipment within the setup. External standards should not be

required unless discrepancies are found. Replace the E]^ TC in the tank by a shorted AN jack. No other changes are necessary. Record the results and file

them in a data book for future reference.

1. Detector vs Lindeck Potentiometer

Zero the detector on the 10-mV range. With the divider switch at

Pot'r and the potentiometer switch at Ej^, adjust the potentiometer to

10.0 mV. The detector should read -10.0 ± 0.1 mV with the set key

' down

.

2. Detector vs Averager

With both keys up, adjust the zero controls on the 150B to give +50.0

div (upper scale) on the 10-mV range. Operate the averager and record

its reading. Readjust the zero controls to give -50.0 div, then

operate the averager and record its reading as before. The total

change in the averager readings should be 100 ± 1 div. If not,

readjust R5 on the averager. (Warning: this is a sensitive control.)

3. Interrange Tests of the Detector

These tests require careful settings but are easy to make and should

suffice to check proper operation. Start as in B-1, on the 10-mV

range. With the set key down, adjust the potentiometer controls to

obtain a 3.0-mV reading on the 150B. Switch to the 3-mV range and

check that the reading is the same after checking the zero with both

keys up. Adjust the potentiometer to obtain a 1.0-mV reading and

switch to the 1-mV range. Continue down to the 30-)tiV range, then set

the RC to obtain a reading of 10.0 fiV on the 30-/xV range of the 150B

with the read key down. Switch to the 10-^iV range and check. Then

proceed down to the l-/iV range.

4. Reversal Compensator (RC)

Check the RC against the detector, as described above in A- 3.

284
APPENDIX 1

Index of Changes in the MJTC Setup

1981

Feb 1981

3/11/81

3/25/81

4/3/81

8/24/81

11/6/81

11/30/81

12/11/81

12/14/81

1982

1/19/82

2/12/82

4/19/82

4/22/82

4/26/82

5/5/82

6/7/82

8/27/82

9/30/82

10/7/82

10/8/82

10/25/82

11/10/82

MJTC comparator set up in A144 Metrology. See diagrams of 2/81.

Ground lead g added and LO-GND link on Keithley 150B opened. See

diagrams of 2/81.

Rj- circuit added and used hereafter for n-compensation.

Grounded black output post at ac amplifier (50 V to ground

with switch on dc)

.

Replaced this ground with 3-/iF capacitor.

Connected two 1000-pF disc capacitors across input leads at tank

and a 1000-pF mica capacitor from each output lead to tank for

radio- frequency- interference bypass

.

Op-amp timer- integrator ("averager") built and installed.

Shielded Lindeck milliammeter (by placing screen over dial) to

eliminate electrostatic pickup.

DVM installed in averager in place of Triplett tester.

Started 10-s integration periods.

Changed ac supply leads to keep amplifier grounded.

Read-2 position installed (first decade of Kelvin-Varley

divider switched out). All tests with two SJTC's performed in

Read-2 position now, unless otherwise noted.

Detector grounding changed. See diagram on p . 2 of data sheets

for this date.

Optimation amplifier replaced by Krohn-Hite Model 7500.

New relays put in ac-dc control box. Bypass capacitor for radio

frequency interference placed outside tank (input leads, only).

Replaced tightly twisted pair of input leads in tank with coaxial

cable. Replaced each TC output (twisted pair) with shielded

twisted pair. Coaxial cable shield and each twisted pair shield

grounded at tank.

Ac input leads changed. See diagram on data sheet for this

date

.

1000-n resistor (in small metal box), formerly across Ei, put in

series in Ej^ line with b lead to form an RC filter with 2-;xF

capacitor across detector leads at EMF comparator. (More detail

is in data book under this date.)

Series resistor for this RC filter removed due to complications

in setting R^-. Installed matching transformer in output of Krohn-Hite amplifier

to avoid errors from dc offset voltage. See data sheets for

10/7/82.

Back-to-back Zener diodes used for TE protection.

50-n General Radio resistor connected in series with each TE

output (and mounted in tank) to avoid adding them when An is

large and Rj- is less than 200 n, the output resistance of the

potentiometer

.

285
12/9/82

12/10/82

1983

4/18/83

5/18/83

Set up Optimation PA 50 T amplifier (modified) and Krohn-Hite MT56R transformer for the ac supply.

Fi TVC connectors (General Radio 874) stabilized by tinned copper

wire wedges

.

Internal 15-/iF series capacitor in Krohn-Hite MT-56R transformer

removed (shorted) to improve low- frequency (20 -Hz) performance.

LC network added to Cohu (Kintel) Model 303B dc voltage standard

(at its terminals) to reduce transients at 50 mA when dc is

switched off.

1985

7/17/85 Changes made in lead connections at input binding posts of tank

to accommodate Wagner ground if needed. Details are in data book

for this date.

APPENDIX 2

AC Pickup in the Detector

The comparator balances out the dc EMF's of the two TE's, but may not balance

out the ac EMF's, since these may not be in the same ratio or phase. They are particularly large at low frequencies, and could cause errors if they overload

the detector. In addition, the 150B has peculiar "windows" of response at or near multiples of the 94-Hz chopper frequency. To reduce these effects, the

150B has a filter in its input. To reduce high-frequency effects, the

comparator has a 2-/iF capacitor across the detector terminals.

In case of doubt, check on possible errors from these normal-mode ac effects

by repeating a test with a higher range on the 150B, which would have less

overload, or by using a different detector.

For further information, see section III-A(6) of NBSIR 84-2903.

286
REFERENCES

F. L. Hermach, "An investigation of the uncertainties of the NBS thermal

voltage and current converters," Nat. Bur. Stand. (US), Rep. NBSIR 84-

2903, Apr. 1985.

F. L. Hermach and E. S. Williams, "Thermal converters for audio- frequency

voltage measurements of high accuracy," IEEE Trans. Instrum. Meas., vol.

IM-15, pp. 260-268, Dec. 1966.

F. L. Hermach and D. R. Flach, "An investigation of mult i junction thermal

converters," IEEE Trans. Instrum. Meas,, vol. IM-25, pp. 524-528, Dec.

1976.

F. L. Hermach, J. R. Kinard, and J. R. Hastings, "Multijunction thermal

converters as the NBS primary ac-dc transfer standards for ac current and

voltage measurements," IEEE Trans. Instrum. Meas., vol. IM-36, pp. 300-

306, June 1987. (This paper was presented at the Conference on Precision

Electromagnetic Measurements (CPEM) held at NBS (Gaithersburg) in June

1986.)

287
Figure A. Diagram of MJTC Setup

K Kintel 303B dc voltage standard

0 Optimation RCD-9 oscillator

A Optimation PA 50 T amplifier

T Krohn-Hite MT-56R matching transformer

M Weston 1 dc milliammeter

S ac-dc relay box

B thermally- lagged metal tank for TC's

C EMF comparator

N detector (usually Keithley 150B)

Rj. Leeds & Northrup ac-dc decade resistor

RC dc reversal compensator

Av electronic "averager"

F fine controls for ac and dc

The cases of K and T are connected to point G on the tank with braid (not

shown). The instruments K, 0, A, and N are grounded through their line cords.

All leads are two-conductor shielded cables unless marked "t" for twisted

pair, "c" for coaxial, or "b" for braid. Bypass capacitors for controlling

radio frequency interference are shown outside the tank. Wiring, bypass

capacitors, etc. inside the tank are not shown; the input to the TC's is

coaxial and their output leads are two-conductor shielded cables with the

shields grounded to the tank.

288
A. Pot'r to E, (E, is "Test")

Rq • Pot'r -vw

Eh

B. Pot'r to Ef, (Eh is "Test")

Read-2

Pot'r

Notes: Eh — Higher EiVIF

E| — Lower EIVIF SK — Set Key

RK — Read Key

D — Divider N — Detector P — Pot'r (200- Q Output Res. is Part of Rf) Rq — 50 - Q Resistor, in Tanl< (to Keep Rr > 200 Q at Balance)

RC — DC Reversal Compensator

Figure C. Simplified Schematics of Comparator

Set

To DC Rev. Comp.

Read

ure B. Diagram of EMF Comparator

289
Output

Rei Re2

DC

Line

'b

cr\jo30-V

DC Supply

Relay Coils

AC DC RDC AC

Figure D. Relay AC-DC Transfer Switch

Inside Tank

Back-to-Back

Zener-Dlode Pair

TE Heater

as Fuse

To TC's

From AC-DC Relay Box

Figure E. Input Connections at Tank

290
FORM NBS-M9

(8-B-eo) U.S. DEPARTMENT OF COMMERCE

SAMPLE DATA SHEET national bureau of standards

TEST NO. DATE - ABCDEFGHIJK a.

zV

f

T

p

tS

K

K

r

irV rc

nu

a

u

a

-

a

°d

jj a

USCOMM-DC 24004-P60 291
Section 17

SCHEDULE FOR RECALIBRATION AND PERIODIC CHECKS

OF NBS THERMAL CONVERTERS

5-10 years Recalibration of reference standards, both thermal voltage converters

(TVC's) and thermal current converters (TCC's), including voltage and

current build-up/build-down and frequency extension.

2-3 years Recalibration of working standards in terms of reference standards.

Routine Checks of working standards according to attached timetables.

TABLE 1. Ranges of low-frequency, thermal voltage converter working

standards, to be intercompared during routine check measurements.

Frequency = 100 kHz.

Voltage Week 1 Week 3

NBS F-7 NBS S/N 302 NBS F-7 NBS S/N 54

1 V 1 V (F1) 1 V (TE-A) 1 V {F1) 1 V (T-2.5)

2 V 2 V (F2) 2 V (TE-B) 2 V {F2) 2 V (T-5)

3 V 3 V (F1) 4 V (TE-C) 3 V (F1) 3 V (R-1, T-2.5)

10 V 10 V (F1) 12 V (R-A, TE-C) 10 V (F1) 10 V (R-2, T-2.5)

20 V 20 V {F2) 20 V (R-B, TE-B) 20 V (F2) 20 V (R-2, T-5)

60 V 60 V (F2) 60 V (R-C, TE-B) 60 V (F2) 60 V (R-3, T-5)

100 V 100 V (F1) 100 V (R-D, TE-A) 100 V (F1) 100 V (R-4, T-2.5)

300 V 300 V (F1) 300 V (R-E, TE-A) 300 V (F1) 300 V (R-5, T-2.5)

600 V 1000 V (F1) 600 V (R-E, TE-B) 1000 V (F1) 1000 V (R-6, T-2.5)

Voltage Week 5

NBS S/N 302 NBS S/N 54

1 V 1 V (TE-A) 1 V (T-2.5)

2 V 2 V (TE-B) 2 V (T-5)

3 V 4 V (TE-C) 3 V (R-1, T-2.5)

10 V 12 V (R-A, TE-C) 10 V (R-2, T-2.5)

20 V 20 V (R-B, TE-B) 20 V (R-2, T-5)

60 V 60 V (R-C, TE-B) 60 V (R-3, T-5)

100 V 100 V (R-D, TE-A) 100 V (R-4, T-2.5)

300 V 300 V (R-E, TE-A) 300 V (R-5, T-2.5)

600 V 600 V (R-E, TE-B) 1000 V (R-6, T-2.5)

The low-frequency TVC sets in Table 1 are identified as NBS F-7, NBS S/N 302, and

NBS S/N 54. NBS F-7 contains two thermoelement modules labeled F1 and F2 and six range

resistors identified according to their voltage ranges. NBS S/N 302 contains three

thermoelements labeled TE-A, TE-B, and TE-C. The range resistors are identified as R-A

through R-E. NBS S/N 54 contains two thermoelements labeled T-2.5 and T-5, and range

resistors labeled R-1 through R-6.

292
TABLE 2. Routine checks for high-frequency thermal voltage

converter standards. Frequency = 1 MHz.

Voltage Standard Check

TVC Standard

0.5 V C1P C0.5

1 V C1P TDO.D-8-1 V

2 V C2P TD0.D-8-3V

3 V C3P TD0.D-8-3V

5 V C6P C5

10 V CIO TD0.D-8-10V

10 V C20 TD0.D-8-10V

50 V C50 TD0.D-8-50V

50 V C100 TD0.D-8-50V

The high-frequency TVC standards and check standards in Table 2 are of the coaxial single-range

type. The standards are labeled with the prefix "C" and check standards are labeled with either

"C" or "TDO".

TABLE 3. Routine checks for working-standard thermal current

converters. Frequency = 50 kHz.

Current Standard Check

TCC Standard

5 mA A3(5 mA)#95 ESW FG (10 mA)

10 mA B10(10 mA)#57 ESW FG (10 mA)

20 mA F20(20 mA)#1 ESW (0.03 FG)

25 mA W30(25 mA)#1 ESW (0.03 FG)

50 mA F50#2 ESW (0.06 FH)

100 mA B1 00#3 ESW (0.1 FG)

200 mA A250#4 ESW (0.2 FH)

500 mA W0.5A#0-2 ESW (0.6 FH)

1 A W1 A#7-1 ESW (1 FG)

2 A W2A#8-3 ESW (2 FH)

3 A W3A#9-2 ESW (3 FG)

5 A W5A#5-1 ESW (6 FH)

10 A W1 0A#0-3 ESW (10 FG)

16 A W20A#1 -2 ESW (20 FH)

The thermal current converter standards in Table 3 are all thermolements with heaters

of high rated current. The check standards consist of a set of current shunts and thermoelements

labeled FG and FH.

293
Section 18

UNITED STATES DEPARTMENT OF COMMERCE

National Bureau of Standards

Gaithersburg, Maryland SOBSS

REPORT OF CALIBRATION

SET OF THERMAL TRANSFER STANDARDS

Manufacturer's Name

Model T, Serial No. 54321

Submitted by

Company ' s Name

City, State, Zip Code

These thermal voltage converters were tested in September 1987 to determine

their ac-dc differences.

Determinations were made of the difference between the alternating voltage (of

practically sine waveform) required for a given output emf and the average of

the two polarities of direct voltage required for the same emf. The observed

differences are generally given to the nearest five parts per million (ppm) in

the table below, where a positive sign signifies that more alternating voltage

was required to produce the same emf. The result at 20 volts and 20 kilohertz

is rounded to the nearest part per million.

Thermal Voltage Voltage Applied

Converter Range Voltage AC-DC Difference (ppm)

Resistor TE Volts Volts 2Hz 5Hz 20Hz 20kHz 50kHz

2 2 2 +20 +10 +5 0 -5

A 1 3 3 +20 +15 0 -10 -10

B 2 20 20 +20 +10 +5 +12 + 15

D 2 200 200 -5 0 + 15

E 1 300 300 +5 +10 +20

E 2 600 600 -5 -5 -15

F 1 1000 1000 +5 -5 -15

Thermal Voltage Voltage Applied

Converter Range Voltage AC-DC Difference (ppm)

Resistor TE Volts Volts lOOkHz 200kHz 500kHz iMHz

2 2 2 -5 +10 +30 +70

A 1 3 3 +5 +15 +25 +40

B 2 20 20 0 -15 -35 +5

D 2 200 200 +80

E 1 300 300 +10

E 2 600 600 +135

F 1 1000 1000 +30

521/Test Number

294
Set of Thermal Transfer Standards

Manufacturer's (brief) Name Model T, Serial No. 54321

Submitting Company's Name

City, State, Zip Code

It is very unlikely that these values of ac-dc difference are in error by more than the amounts listed in the table below. These figures include allowances

for both the random and systematic uncertainties of the calibration process.

No allowance has been made for the long-term stability of these standards.

Applied

Voltage Uncertainty (ppm of applied voltage)

Volts 2Hz 5Hz 20Hz 20kHz 50kHz lOOkHz 200kHz 500kHz IMHz

> 100 20 20 30 50

5-100 200 100 20 15 25 40 100 100 200 < 5 200 100 20 20 30 50 100 100 200

The uncertainty for the B,2 combination at 20 volts and 20 kilohertz is 10 ppm

of the applied voltage. This lower uncertainty was achieved after additional

measurements were made at this test point (by prior arrangement) with

different standards and a more precise comparator system. A general

discussion of the uncertainties is presented in "An Investigation of the

Uncertainties of the NBS Thermal Voltage and Current Converters" by F. L. Hermach (NBS Internal Report NBSIR 84-2903).

For the Director

National Measurement Laboratory

Norman B. Belecki, Physicist

Electricity Division

Center for Basic Standards

Test No.: 521/Test Number

Reference: Purchase Order Number

Date: September 30, 1987

295
NBS-n4A (REV. 2.8C)

1 PUBLICATION OR 2. Performing Orgdn* Report No, im Publicdtion Dste BIBLIOGRAPHIC DATA

REPORT NO.

SHEET (See I'nstruct/ons) NIST/SP-250/27 May 1989

4. TITLE AND SUBTITLE

NIST Measurement Services:

AC-DC Difference Calibrations

5. AUTHOR(S)

Joseph R. Klnard, John R. Hastings, Thomas E. Lipe, and Clifton B. Chllders

6. PERFORMING ORGANIZATION (If joint or other than NBS. see instructions) 7. Contract/Grant No.

NATIONAL INSTrrUTE OF STANDARDS AND TECHNOLOGY

(formerly NATIONAL BUREAU OF STANDARDS) 8. Type of Report & Period Covered

U.S. DEPARTMENT OF COMMERCE

GAfTHERSBURG, MD 208^ Final

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street, City. State, ZIP)

Same as item 6

10. SUPPLEMENTARY NOTES

Library of Congress Catalog Card Number: 89-600736

I I Document describes a computer program; SF-185, FIPS Software Summary, Is attached.

11. ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here)

This publication collects and summarizes the specialized information needed to

operate the ac-dc difference laboratory and calibration service atiMIST

(Gaithersburg) . It also serves as a convenient reference source for the users

of this calibration service and other interested people by documenting the

service and its underlying background in considerable detail. It contains

the following: an annotated table of contents, a topical index, and a glossary

of common ac-dc acronyms; an overview of the service; selected published

papers; instructions for the operation of the comparator systems; a schedule

for the recalibration and periodic checks of the [\|IST thermal converters; and

a sample report of calibration.

12. KEY WORDS (S/x to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

ac-dc difference; ac-dc transfer; calibration; comparator; current; measurement;

standards; thermal converter; uncertainty; voltage

13. AVAILABILITY

Pxl Unlimited

I I For Official Distribution, Do Not Release to NTIS

[X] Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C.

20402.

Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

308

15. Price

^r U.S. GOVERNMENT PRINTING OFFICE: 1989— 242-311/ 0^*060 USCOMM-DC «043-PeO
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