A11107 nA04Q

7 . PUBLICATIONS

NBS Special Publication

Measurement

of Electrical Quantities

in Pulse Power Systems

100

• U57

No, 623

1982

c. 2

NBS
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k I /) / i *

* i^St rjtjt NATIONAL BVR»>

Measurement of Electrical Quantities

in Pulse Power Systems

Proceedings of the Workshop on Measurement of Electrical Quantities in Pulse Power

Systems held at the National Bureau of Standards, Souider, Colorado, March 2-4, 1981

JUL 1 9 1982

Dot &cc~dj,c

c: >-

R. H. McKnight and

R. E. Hebner, Jr., Editors

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

Washington, DC 20234

Sponsored by:

The Air Force Office of Scientific Research; the Defense Advanced

Research Project Agency, Directed Energy Office; the Department

of Energy, Office of Inertial Fusion; the Naval Surface Weapons

Center; the U.S. Army, Harry Diamond Laboratories; and the

National Bureau of Standards, Center for Electronics and

Electrical Engineering.

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued June 1 982
Library of Congress Catalog Card Number: 82-600535

National Bureau of Standards Special Publication 628

Nat. Bur. Stand. (U.S.). Spec. Publ. 628, 420 pages, (June 1982)

CODEN: XNBSAV

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For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402

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Foreword

Pulse power systems find application in a wide range of industrial,

educational, and government organizations. As these systems are extended to higher power levels and to more critical applications,

there is an increasing requirement for accurate electrical measurements.

These measurements are necessary in the design, the operation and the

procurement testing of pulse power systems.

To bring together active researchers in the pulse power field,

a workshop on Measurement of Electrical Quantities in Pulse Power

Systems was organized and held at NBS-Boulder on March 2-4, 1981.

These proceedings are a record of that workshop.

The purposes of the workshop were (1) to disseminate information

on modern measurement techniques, (2) define the state-of-the-art of

measurement of electrical quantities in pulse power systems, and

(3) identify specific areas in which improvements in measurements are required. The attendees were representatives from industry, university,

and government.

The workshop was sponsored by the Air Force Office of Scientific

Research; the Defense Advanced Research Project Agency; the Department

of Energy; the Naval Surface Weapons Center; Harry Diamond Laboratories;

and the National Bureau of Standards, Center for Electronics and

Electrical Engineering. Without this sponsorship, the workshop and

publication of these proceedings would not have been possible. An

acknowledgment must also go to the organizing committee whose efforts

contributed significantly to the overall success of the workshop.

ill
Preface

The purpose of these proceedings is to provide a record of the

workshop on Measurement of Electrical Quantities in Pulse Power

Systems. Papers to be presented at the meeting were selected on the

basis of a review by the organizing committee of contributed abstracts.

Some of the papers presented at the workshop are included here by

abstract only because the final manuscripts were not submitted by

the authors.

The papers were organized and presented in four topical areas:

(1) Voltage Measurements, (2) Power and Energy Measurements, (3) Current

Measurements, and (4) Data Acquisition.

Open discussion periods were held at the end of each technical

session. These discussions were recorded, transcribed, and returned

to the discussers for editing. What appears in the proceedings are those edited comments. A number of figures and references have been

added for clarity.

The papers have been reproduced as received from the author(s).

During the discussions and technical presentations, trade names were

used. In general, they were included to specify adequately the

experimental procedures used. Such identification does not imply

that the materials or equipment identified are necessarily the best

available for the purpose.

Ronald H. McKni ght Robert E. Hebner, Jr.

Editors

iv
Abstract

The Workshop on Measurement of Electrical Quantities in Pulse

Power Systems addressed measurements in pulse systems having the

characteristics of pulse duration less than a millisecond, system

voltages greater than 10 kilovolts, and system currents greater than

10 kiloamperes. The presented papers were divided into four

categories: voltage measurements, current measurements, power and

energy measurements, and data acquisition. Included are discussions

of applications of conventional measurement techniques and

state-of-the-art systems.

Key words: current measurements; electrical measurements; electromagnetic

pulse; fusion; nuclear effects simulation; particle beam technology;

pulse power; transients; voltage measurements.

v
Organizing Committee

The following individuals were members of the organizing committee

and acted as reviewers for abstracts submitted -4 n response to the

announcement of the workshop.

J. R. Bayless - Defense Advanced Research Projects Agency

M. Di Capua - Physics International

A. H. Guenther -Air Force Weapons Laboratory

R. L. Gullickson - Defense Nuclear Agency

D. G. Gritton - Lawrence Livermore Laboratory

R. E. Hebner, Jr. - National Bureau of Standards

M. Kristiansen - Texas Tech University

R. Lamb - Harry Diamond Laboratories

R. A. Lawton - National Bureau of Standards

R. H. McKnight - National Bureau of Standards

E. Nolting - Naval Surface Weapons Center

D. Pel linen - GCA Corporation

K. R. Prestwich - Sandia Laboratories

C. E. Rossi - Department of Energy

W. J. Sarjeant - Los Alamos Scientific Laboratory

J. Shannon - Maxwell Laboratories

J. D. Shipman, Jr. - Sachs/Freeman Associates

Special Acknowledgments

Several individuals by virtue of their extra efforts contributed

significantly to the success of the workshop. Ken Prestwich was instrumental in securing keynote speakers for the various technical

sessions. Bob Lawton's assistance in coordinating arrangements for

facilities at the NBS-Boulder complex was most helpful. The conference

secretary, Sandra Kelley, provided invaluable support throughout

the process of organizing, arranging, and completing the workshop.

vi
Session A - Voltage Measurements

Electro-optical Pulsed Voltage Measurements,

J. E. Thompson

High-Voltage Probe for Liquid Immersion,

N. W. Harris (

Experimental Comparison of Step Response and Ramp Response

Measurements in Free-Standing Dividers,

R. E. Hebner, Jr.

A Comparison of Three Different Designs of Resistor Divider,

T. R. McComb, M.M.C. Collins, and W. J. Sarjeant

A 100-kV, 2-ns-Ri set i me, dc-Coupled Probe,

J. Power, W. Nunnally, and D. Young

Discussion 1

Calibration of Capacitive Voltage Probes in Water-Dielectric,

High-Power Pulse Generators,

M. Wilkinson and E. Chu

Measurement of Transient Potentials in Coaxial Transmission

Lines Using Coaxial Dividers,

N. Fujimoto, S. A. Boggs, and R. C. Madge

A Voltage Monitor for Magnetical ly- Insul ated Transmission

Lines,

R . W. Stinnett

Measurement of Magnetically Insulated Low Voltage Using a Thomson Parabola Charged Particle Analyzer,

T. D. Stanley and R. W. Stinnett

Discussion 2

Session B - Power and Energy Measurements

Ion Current and Voltage Determinations by Nuclear Techniques,

F. C. Young

Electrical Measurement Techniques Used at the Casino Facility,

E. Nolting, R. Martin, and M. Ruppalt

vi i
Indirect Measurements by Computer Simulation,

R. A. Hill 133

Calibration of the Blackjack 5 Pulse Generator Output Power,

R. Richardson, E. Chu, W. Clark, J. Shannon, and M. Wilkinson 150

Discussion 3 165

Session C - Current Measurements

Rogowski Coils, Fluxmeters, and Resistors for Pulsed Current

Measurements,

M. S. Di Capua 175

Precise Measurement of Current in Pulsed Power Systems,

M. E. Wilmer and P. A. Pearson 194

Low- Inductance Shunts for Measuring Large Pulsed Currents,

W. F. Praeg 204

Measuring Fast Pulse Currents Using Low Inductance Current

Viewing Resistors and DI/DT Probes,

C. A. Muehlenweg and G. McDuff 217

Wide-Frequency Range Current Transformers and Application to

Pulsed Power Systems,

J. M. Anderson 233

Discussion 4 244

Data Acquisition and Processing Techniques,

R. E. Kolibas, P. A. Corbiere, and J. J. Mori arty 248

A Technique for Measuring Beam Current Density in the Antares

Electron Gun (abstract),

W. R. Scarlett 256

Thomson Spectrometer Measurement of Heavy Ion Beams Produced

by a Pulse Powered Plasma Focus Device,

M. J. Rhee 257

Current Measuring Diagnostic Techniques for High DI/DT

Particle Beam Accelerators (abstract),

D. H. McDaniel 266

Proton Current Measurements Using the Prompt Gamma Ray

Diagnostic Technique,

R. J. Leeper, E.J.T. Burns, D. L. Johnson, and W. M. McMurtry 267

vlil
The Measurement of Pulsed Electric Currents by the

Faraday Effect,

J. Katzenstein, W. Caton, and G. M. Wilkinson

Cavfty Current Monitors,

J. Shannon, E. Chu, R. Richardson, M. Wilkinson,

and C. Trivel piece

Discussion 5

Session D - Data Acquisition

High Voltage Monitoring and Control Through Fiber Optics,

J. G. Stewart, Jr. and W. A. Petty

Remote Command Data Link Provides Enhanced Simulator

Performance in High EMP Environment,

S. Lyons

A Fiber Optic Monitoring System for Antares Pulse Power

System (abstract)

,

M. E. Thuot and W. R. Scarlett

Discussion 6

Data Recording Techniques for the Sandia Particle Beam

Fusion Accelerator,

W. B. Boyer and E. L. Neau

An Evaluation of Digital Recording Equipment and Numerical

Correction Techniques in Impulse Measurement,

R. Malewski, T. R. McComb, and M.M.C. Collins

Analysis of a Power System Transient Recording Laboratory,

R. E. Poliner, amd T. J. Reed

Application Problems Using Instrumentation Amplifiers in the

Pulse Power Environment,

E. E. Cunningham

Discussion 7

Digital Correction of Cable Attenuation Losses,

C. Trivelpiece, R. Richardson, J. Shannon, and J. B. Smith

Precision Picosecond-Microsecond Electromagnetic Waveform

Measurements at NBS,

R. A. Lawton

Discussion 8

ix

Session A - Voltage Measurements

ELECTRO-OPTICAL PULSED VOLTAGE MEASUREMENTS

J. E. Thompson

College of Engineering

University of South Carolina

Columbia, SC 29208

Pulsed voltages are typically measured using resistive or capacitive voltage dividers. Problems associated with these types of measurements include the effects of stray capacitance, oscillations, noise

pickup, and ground loops. Many of these problems can be reduced and

measurement frequency response improved by using electro-optical

measurement techniques. The most useful optical techniques for voltage

measurement are the Kerr and Pockels effects. The relevant characteristics of these two effects are discussed together with optical techniques useful for utilizing the Kerr effect to measure pulsed voltages.

Frequency response, accuracy and calibration of the measurement techniques are discussed in the context of high voltage measurements.

Specific electro-optical measurement techniques and devices including

cross polarizer analyzers, interferometric measurements, matched

impedance Kerr cells, a sampling transient analyzer, and a traveling

wave measurement technique are discussed.

Key Words: accuracy, calibration, electro-optical measurements,

frequency response, interferometric measurements, Kerr effect, Pockels

effect, polarization.

1 . Introduction

Pulsed, high voltage measurements are typically made using resistive or capacitive voltage dividers. The frequency response , and

hence the fastest measureable risetime, are limited by stray capacitance

and inductance associated with the divider circuitry. Typical, minimum

measureable risetime, obtainable with a carefully designed divider, is

greater than 1 nanosecond. The signal fidelity and signal to noise

ratio, obtainable with lumped parameter divider techniques, is also

adversely effected by ground loops and electromagnetic interference.

Improved pulsed voltage measurements, with respect to time response,

noise immunity, and sensitivity can be obtained using electro-optical

measurement techniques. The general arrangement necessary for electro1
optical measurements is shown in Figure 1 and consists of a polarized

light source, an electro-optical device, a polarization analyzer, and

a light detector.

»<t) ltl

Probing light

Electro-optical device

Figure 1. Electro-optical Voltage Measurement.

Voltage v(t) to be measured, is applied across the electro-optical device. Polarized light is passed through the device while the voltage

is applied. The light polarization state is modified, due to the relevant electro-optical effect (usually the Kerr or Pockels effect), in a manner dependent upon v(t). The modified polarization state of the

light is passed through an analyzer, which produces a relationship between polarization state and light intensity. The output light intensity, which is now related to v(t), is detected and recorded.

The general electro-optical technique described produces a light

intensity which is dependent upon v(t). The measurement technique has

several advantages compared to more conventional, resistive or capacitive divider techniques. The electro-optical measuring device, shown

in Figure 1, is connected directly, in parallel, with v(t). No divider

circuitry is required. This reduces oscillations and risetime degradation associated with stray inductance and capacitance. The frequency

response, for a carefully designed system, is limited by the capacitance of the electro-optical device and the characteristic relaxation

time of the measurement medium used. Risetimes of less than one nano- second have been directly observed using electro-optical techniques.

System risetime responses as fast as tens of picoseconds have been projected as a realistic measurement capability.

The light source and detector are not physically connected to v(t).

This permits rather complete ground loop and electro-magnetic interference isolation. This isolation capability implies that improved

signal to noise ratios and signal fidelity can be produced. Additionally, as will be shown in this report, the sensitivity of electrooptical measurement techniques improves at higher voltages. This makes

it possible to measure voltage amplitudes with precision on the order

of 1%.

2
This paper will describe the electro-optical effects most useful

for pulsed voltage measurements and optical techniques for producing

light intensity variations which can be related to v(t). Practical

measurement systems and typical results will also be presented.

2. Electro-optical Effects

Several electro-optical effects exist which can be used to measure

voltage. The most useful of these are the Kerr and Pockels effects.

The polarization state of light, which passes through electrically

stressed a Kerr or Pockels effect medium, is modified in a manner dependent upon the value of the voltage applied. The Pockels effect has

been used to measure voltages in the range of tens of kilovol ts [1]. This

effect utilizes relatively expensive crystals such as KDP, KD*P, and

LiNb03 and is limited to voltage measurements in this range, unless

voltage division techniques are utilized. The Kerr effect utilizes

electro-optical liquids, such as nitrobenzene, and can be used, directly,

at voltage levels of several hundred kilovolts. The Kerr effect is

therefore most appropriate for pulsed, high field measurements.

The general arrangement for electro-optical measurement has been

shown in Figure 1. The electro-optical device shown consists, for high

voltage Kerr effect measurements, of an electrode structure filled with

Kerr effect liquid. A simple arrangement for the "Kerr cell", electrooptical device is shown in Figure 2 and consists of 2 parallel plate

electrodes separated by a volume filled with Kerr effect fluid.

+ v(t) p

Figure 2. Kerr Electro-optical Effect.
Linearly polarized light, having polarization_components parallel, PJt , and perpendicular, Px , to the applied field, E, is passed through the

Kerr cell. A phase shift <$> is introduced, because of the Kerr effect,

between the orthogonal light components. This introduced phase is re- lated to the local value of the electric field at a point x, y and time

t according to,

<Kx,y,t) = 2ttKL E(x,y,t) • E(x,y,t), (1)

where K is constant(the Kerr coefficient) and L is the length of the

Kerr cell in the direction of light propagation. The electric field,

E, can be assumed, for short pulses having a duration of less than 100

us, to be uniform in the central electrode region and equal to v(t)/d,

where d is the electrode separation. The applied voltage is therefore

given, in terms of <j>, by

v(t) =

This implies that measurement of 4>(t), in the uniform field region, will

determine v(t). It should be noted that his relationship neglects the

effects of fringing fields at the ends of the Kerr cell. These effects

can be neglected for most Kerr cells of practical lengths. A quantitative analysis concerning the effects of fringing is given by Thatcher [2]

3. Measurement of <f>, Polarizing Light Intensity Variations Indicative

of v(t)

The applied voltage v(t) is related to <p{t) . The quantity <j>(t) can

be measured in several ways. The most common technique involves the use of crossed polarizers. An alternate technique, which has advantages for

certain applications, uses a polarization interferometer. This section

will present the general measurement principles and will discuss waveform shape, interpretation, calibration, errors, and temporal response

considerations.

3.1 Cross Polarizer Measurements

The cross polarizer analyzer arrangement necessary for measurinq

<j)(t) is shown in Fiqure 3 and consists of linear polarizers P450 and

P_45° and quarter wave plates Qq and Qgg, placed before and after the

Kerr cell. The polarizer P45 linearly polarizes the incoming light into

components Ph and Pj_, oriented parallel and perpendicularly to the applied field E. Polarizer P_^^ is oriented orthogonally to P45

such that no light will pass through P-45, if the Kerr cell is isotropic.

The Kerr liquid, however, becomes anisotropic as voltage is applied.

It can be shown that the intensity of the transmitted light is given by,

*(t)

2 KLd

1/2

(2)

4
I(x,y,t) = I sin

2 [*(x,y,t)/2], (3)

where I m

is a constant and corresponds to the maximum transmitted light

i ntensity

Kerr fluid

I

1 X/4

45

1

1

1

! X/4

-45

Figure 3. Cross Polarizer, Kerr Electro-Optical Measurements.

The quarter wave plates CL and Qgo convert the linearly polarized

light to circularly polarized light and back again to linearly polarized

light. Their purpose is to make the transmitted ligth intensity independent of the direction of E. This is described in detail in reference

[3].

The spatial variation implied by Eq. 1 is shown in Figure 4.

The photograph shows the shadow of the electrodes and dark curves, or fringes, corresponding to the minima of Eq. 3. The electric field at any

location x,y can be determined from the data of Figure 4 by using Eq. 3

along each of the fringes.

The electric field at one point, located in a region of uniform field,

can be determined by measuring the intensity transmitted at that point,

as a function of time. A typical applied voltage, to be electro-optical

ly measured, and the corresponding, transmitted intensity is depicted in

Figure 5. The voltage rises to a flat top and then falls to zero. The

output intensity, in terms of v(t), is given, using Eqs. 2 and 3 as,

Kt) = I m

sin'

irKt/(t)

(4)

5

An actual I(t) is shown in Fiqure 6 together with v(t) as measured

using a resistive divider. It can be seen that I(t) experiences maxima

and minima as the voltage increases and decreases in time. The I(t) of

Figures 5 and 6 can be related to v(t), by solving Eq. 4 for v(t),

v ( t) = 4: *^

This inversion can be performed, for all t, for well behaved waveforms

and be can be automated for predictable waveshapes such as pulses. The

inversion, however, becomes somewhat ambiguous for complicated temporal

waveshapes, such as damped sinusoids.

Resistive divider

Kerr fringes

Figure 6. Cross Polarizer Kerr Electro-optical Effect

Voltage Measurements (Hebner,et al [1 ]).

Kt)

(5)

7
The v(t) can also be determined from the I(t) shown in Figure 6 by

noting that I(t) experiences minima when,

^KLv^t) = n^ ; n = 0,1 ,2,,

.

,2

(6)

The number of minima or "fringes" can be counted, as a function of time,

to yield,

v(t) = d n(t)

KL

(7)

It should be noted that even though Eq. 7 was derived assuming inteqer

values of n, partial fringe shifts can be measured such that the value

of v(t) at times not corresponding to minima can also be determined.

Partial fringe shifts is small as 1/4 are easily measureable. The re- sults shown in Eq. 7, therefore, imply that v(t) can be determined by

counting the number of fringes produced, as a function of time, n(t).

This counting can also be automated.

3.1.1 Measurement Accuracy

Equation 1 shows that the voltage is proportional to the square root

of the number of fringes observed. This implies that the change in n,

An, for a change in v, av, is given by,

An = 2KL vav . (8) ~7

This, in turn, implies that An, for a given av, increases as the voltage

increases. The sensitivity of the measured n to a change in v therefore

increases at higher voltage levels. As an example, assuming v = lOOkV,

L = 0.1m, d = 0.002m and a minimum detectable An of 0.25, implies a minimum, measureable av of 154V. The resolution implied is therefore 154/

lOOkV = 0.15%. The resolution and sensitivity to peak voltage values

therefore improves substantially at higher voltages. This of course is

a positive attribute for high voltaae pulse measurements and is an advantage of the electro-optical technique.

3.1.2 Calibration

The Kerr effect measurement can be calibrated by applying a voltage

value to produce the second minimum, corresponding to n = 1 of Eq. 6.

This voltage value, V], is termed the cell constant and, from Eq. 7, is

given by,

3
1/2

(9)

The voltage V-| is typically applied using a calibrated source or measured

using a calibrated, resistive divider. Voltage values can therefore be

determined, after calibration, in terms of V] according to,

v(t) = V^t)]*. (10)

3.1.3 Measurement Errors

Errors, associated with the electro-optical measurement of v(t),

arise due to errors in measurement of V] and errors involved in the measurement of <J>. Using Egs. 2 and 10, it can be shown that the percentage

error in measuring v, av/v, in terms of measurement errors in Vi , aV-|/Vi , and <j>, Acf>/<j), is given by

av_ _ + aV

-J + 1_ Acj)_ . (11

)

v - -w- - F <j> v

l

The values of AV/V-j and 1 /2a<j>/4> are typically 1% each [4]. This implies

that the error associated with measurement of v is on the order of a few

percent.

3.1.4 Temporal Response

The temporal response of the electro-optical measurement technique

is determined by the responses of the Kerr cell, the Kerr cell leads,

the light detector and recorder. The frequency response of the Kerr cell

is determined by the capacitance of the Kerr cell, for lumped Kerr cell

measurements, and the intrinsic relaxation time of the Kerr liquid for

distributed Kerr cell measurements. The Kerr cell liquid is typically

nitrobenzene, which has a high relative permittivity of approximately 36.

The resulting capacitance, C, of a nitrobenzene Kerr cell is, therefore,

on the order of tens of picofarads. The risetime capability of lumped

Kerr cells of this type is determined primarily by the Kerr cell capacitance and the circuitry used to connect the Kerr cell to the voltage

being measured. The risetime of Kerr cells driven with a transmission

line is given by Z~C, where Zq is the characteristic impedance of the

line, and is on the order of a few nanoseconds for 50fi cable. Lower

impedance, coaxial or parallel plate, transmission line connections re- sult in faster risetimes. It should therefore be clear, that for lumped

Kerr cells, connected to the voltage to be measured with a transmission

line, low capacitance and low driving impedance are necessary to accomplish less than nanosecond responses. It should be noted that for proper

I

KL

9
operation, the Kerr cell should terminate the line in its characteristic

impedance. Bergman has satisfactorily used a Kerr cell and transmission

line connections to measure voltage risetimes on the order of Ins [5].

The connection between the lumped Kerr cell and v(t) can be made

with inductive leads. The measurement frequency response is, in this

case, limited by the resonant frequency of the leads and the Kerr cell

capacitance. This resonant frequency is on the order of 1 x lO^HZ which

implies a risetime capability of 3.5ns [6].

The frequency response of Kerr effect is intrinsically limited by

the natural relaxation time of the liquid used. This time is on the

order of 50ps for nitrobenzene, as little as 2ps for CSp, and is the

ultimate limit to risetimes, measureable using the Kerr effect [7].. This

limit is, however, applicable only if capacitance and inductive effects

are negligible; as in the case of a travelling wave measurement system

to be discussed in a later section of this paper.

Equations 7 and 8 imply that the sensitivity of the measurement

technique increases at higher field values due to the fact that more

fringes are created, per field increment, at high voltages. This implies

that a large number, N, of fringes can be created during the measured

pulse risetime.

This is illustrated in the detected output shown in Figures 5 and 6.

This further implies that the frequency response of the light detector

used must be on the order of N times the frequency response implied by

the measured pulse risetime. There, therefore, must be a compromise

between sensitivity/accuracy, or equivalently number of fringes, and

measureable risetime capability of the electro-optical system.

i

The frequency response of available photo detectors, capable of

driving a low impedance oscilloscope or coaxial cable is limited. Improved frequency response capability can be attained by using an image

converter camera as the detector. This is accomplished by streaking thr

fringe pattern observed as I 0u f

A typical fringe pattern,

obtained using the arrangement of Figure 3 and a pulsed laser, is shown

in Figure 7. The observed intensity distribution is again given by Eq.

3. Fringes, corresponding to the minima of Eq. 3, are seen to occur in

space and are indicative of the fact that the electric field distribution, in the region shown, is spatilaly nonuniform. Few fringes are observed in the indicated, central region, where the field is most uniform. The intensity transmitted through the slit will be alternately

bright and dark as the electric field increases from a zero to a peak

value. This temporal change in intensity can be detected using an image

converter camera to streak the transmitted frinqe pattern. Results of

this type are shown in Figure 7. A uniform field in the slit region

would produce vertical alternating bright and dark bands of the field

increases from 0, at Oys, to a peak value, at 4.8ys. The observed

bending of the fringes is due to the fact that the field, in the slit

region, is spatially nonuniform. The fringes are observed to remain

relatively straight in the indicated central region. These unbent

10
fringes can be used to determine the temporal performance of the electric

field, E(t), in the central region and hence the applied voltage, v(t).

This is done by counting the number of fringes, n(t), as a function of

time, and using Eq. 10.

Figure 7. Streaked Cross Polarizer Kerr Electro-optical

Voltage Measurement (Hebner, et al [ ]).

The image converter camera, used as a detector, produces measurement data shown in Figure 7, using the crossed polarizer arrangement of

Figure 3. This compares to analogous measurement data, obtained using

a photodetector, shown in Figure 6. The advantages of the image con- verter measurement is the fast writing speed available. Using writing

speeds of O.lmm/ps it is possible to discern a relatively large number

of fringes, during the pulse risetime, in a short time period. Good

sensitivity, accuracy, and superior risetime capability are therefore

achievable.

The temporal response discussions have addressed the shortest ri se- time or time variation which can be measured. It should, however, be

noted that there is a limit to the longest pulse which can be effectively

measured. Long pulses cause volume space charge accumulation in the Kerr

measurement fluid. This charge distorts the inter-electro electric field

and complicates measurement results [8]. Volume space charge effects

are not important for nanosecond duration pulses. The pulse duration at

which the effects do become important depends on the resistivity of the

Kerr fluid. Higher resistivity values are required for longer pulses.

Reasonable care can produce nitrobenzene bulk resistivity in the ranae

of ICPft-cm. Resistivities, on this order, permit the measurement of

pulse lengths up to several hundred microseconds.

3.2 Interferometric Phase Measurements

The phase difference <j>, introduced due to the Kerr effect, can be

measured using the crossed polarizer arranaement of Figure 3. The rela11
tionship between <|>, E,v, and n have been presented in this paper. This

phase measurement technique produces a frinqe pattern in time, as shown

in Figure 6 or a fringe pattern in space, as shown in Figure 4, which

can be streaked in time to hi el d measurement data, as shown in Figure 7.

The temporal data of Figures 6 and 7 are interpretable for simple waveshapes such as flat pulses with little oscillatory overshoot or undershoot. However, electro-optical results produced by complicated waveforms, such as damped sinusoids, or pulses with oscillatory components,

are difficult to interpret. This arises due to the fact that the relationship between v(t) and I(t) as given by Eq. 4, is nonlinear and multivalued. Certain of the ambiguities, associated with the relationship

between v and I ,can be eliminated using two probing wavelengths, simultaneously, to produce two separate sets of measurement data similar to

that shown in Figure 6 or by using a Kerr cell having two probed lengths [9]

An alternate technique for measuring <j>, which can produce more

interpretable temporal voltage measurements and sensitive spatial field

measurement, utilizes the polarization analyzer shown in Figure 8. Linearly polarized light, having components, parallel P„ , and perpendicular, 5. » to the applied field are passed through the Kerr fluid, as shown in Figure 2. The light, having a phase shift <}>, between the orthogonal components, then enters the analyzer. The analyzer produces output beams P and P in the two interferometer paths, using the polarizers

and halfwave plate shown. These two beams are then recombined, using

the beam splitter shown, to produce a fringe pattern indicative of <j>. The detailed operation of the analyzer, as used for field measurements

is given in reference [10].Typical results obtained using a pulsed laser

are shown in Figure 9. The interferogram shows the shadow of the two

excitation electrodes and fringes formed between the electrodes in the

nitrobenzene filled region. The position of the fringes is related to

the local field value. Interference patterns of the type shown in Figure

9 are useful for measuring field spatial distributions, at a specific

time. These fringes can also be streaked in time, in a manner analogous

to that used to obtain the results of Figure 7, by positioning a vertical

slit, as described previously. Results obtained in this manner are shown in Figure 10. Figure 1 0 (aTshows , as~the~upper trace, the v(t) to

be measured and, as the lower trace, the measurement of I(t) using the

crossed polarizer technique of Figure 3. It should be clear thatsome

interpretation difficulties exist for relating I(t) to v(t) in this case. Similar measurement results, obtained using the analyzer of Figure 8 and

fringes, as shown in Figure 9, streaked in time, are shown in Figure

10(b). It can be observed that each streaked fringe^ has a temporal shape

which is clearly related to the excitation waveshape. It can be shown

that the measured field values are given by,

1/2

E(t) = Ax(t)[

6xKL

(12)

where 6x is the spacing between undiviated fringes (corresponding to zero

field) and Ax(t) is the measured deviation of the fringes [9].. The quantity

12
•e- o

<u

CLE

03<D CO

r- CD CL Uc

CT> CD > C S-u-

•r- <d -M

s_ t_ o o ai 'r- o +-> -o

DC t—i -r0)>

118 cu

fO

:x: CL

i-<uM

"r— J_

(tt 'o

a.

atu

icuN

•r—i-

(13 *o

CD

CL

</> E

fO

a> ca \

in

4->£=CDE

CL) Srs

td

OJ

5-Oo

sai

•)-> cuEo

OJ

4-JNLo

00e

=3

C7>

13

i_ ->

aito CD f—

_i
Ax(t) can be directly measured from the interferogram to yield the temporal performance of E(t). The voltage v(t) is clearly related to E(t)

for uniform field electrode geometries.

Fringing fi

Central field regi

Figure 9. Typical Interferogram of Field Distribution Between Parallel

Electrodes HOI.

5kV/cm

20 ys/cm

(a) Voltage Divider and Cross

Polarizer

40 ys/cm

(fe) Interferometric

Figure 10. Cross Polarizer and Interferometric Measurements CO]

14
The streaked interferometric measurments , illustrated in Figure 10(b),

also have sensitivity that increases with v. The results can be viewed

somewhat directly, similar to an oscilloscope trace . There is therefore

less ambiguity in relating I(t) to v(t). It should, however, be clear

that additional expense and complication is associated with this technique

primarily due to the need for an image converter camera. The technique

if however attractive for certain applications where complicated, fast

pulses are to be measured. Results to date have been produced, utilizing this technique, with subnanosecond resolution[ll]. It should however

be possible to extend this to 10ps resolution, using the fast streak

cameras (lOps/mm) currently available.

4. Typical Pulse Measurement Systems

The general principles assocaited with electro-optical voltage

measurements have been discussed in the previous sections. Several

practical systems, used by various researchers, will be described in this

section. The first utilizes on impedance matched Kerr cell to improve

the risetime and a sampling pulsed laser. The second uses a pulsed

laser and optical delay lines to determine fast risetimes. The last

technique utilizes a coaxial geometry appropriate to very fast traveling wave measurements.

4.1 Impedance Matched Kerr Cell

Voltage measuring Kerr cells are primarily capacitive. The frequency response of these cells is, therefore, limited by the time required

to charge this capacitance or by resonant oscillations between the capacitance and lead inductance.

Bergman and Kolleogy have used impedance matched Kerr cell to measure fast voltage behavior associated with a discharge in N2 gas[5]. The

measurement Kerr cell is shown in Figure 11 and consists of a parallel

plate transmission line which is fed by a coaxial cable connected to the

voltage being measured. The voltage pulse propagates from the source,

through the cable and the Kerr cell>and out another load coaxial line. The

dimensions of the nitrobenzene filled Kerr cell are chosen to produce a parallel plate transmission line having the same characteristic impedance

as the driving and load cables. No reflections, therefore, occur. The

risetime of the response is therefore not limited by the capacitive

charging, 1 ine impedance, or by oscillations due to the Kerr cell capacitance and lead inductance.

The cell shown in Figure 11 has been used in - conjunction with a pulsed dye laser to permit voltage measurements at discrete Ins times.

The temporal resolution capability is, therefore, on the order of Ins.

4.2 .Optical Sampling Transient Analyzer

A nitrobenzene filled transmission line has been used by Davies and

Nelson to measure the risetimes of fast pulses [12]. This system is shown

in Figure 12. The voltage to be measured propagates down the line. Light

15
from a pulsed laser travels through fiber optic cable and enters the line,

through linear polarizers, passes through the nitrobenzene, and exits on the other side of the line, through crossed polarizers. The light entry

and exit parts are displaced along the line as shown. The Kerr effect at

these entry ports, therefore, measures the pulse voltage at various times,

which can be related to the risetime and propagation velocity of the wave.

A larger or smaller time interval of the excitation can be measured, with

a given port separation, by using fiber optical cables of different

lengths. The frequency response of this system is on the order of 1 GHz.

Electrical Filling Electrical

transient holes termination

Opti caj

Figure 11.

Mounting ^ holes

Schematic Drawing of the Kerr Cell with Electrical

Connections Indicated [ 5].

Detector and

electronics

Crossed polarizers

on this side

Optical delay

1 i nes

Pulsed laser

Figure 12. Optical Sampling Transient Analyzer (Davies and Nelson [121).

16
4.3 Coaxial Field Measurements

Wunsch, Sanchez, and Green have used the Kerr effect to measure the

electric fields within pulse excited coaxial cyl inders[7] . The experimental

arrangement is shown in Figure 13 and consists of a water filled coaxial

transmission line with optical input and output ports. The electromagnetic pulse, to be measured, propagates down the line. Light is directed

through a polarizer into the area between the coaxial cylinders and propagates in the same direction as the pulse. The light beam is positioned

nearer the center conductor to be in a higher field region to improve mea- surement sensitivity. The polarization state of the light changes (due

to the Kerr effect of water) as the pulse fills the volume between the

two mirrors. The risetime of the optical response is therefore dependent

upon the risetime of the pulse, the propagation time between the mirrors,

and the response time of the Kerr fluid. Megavolt signals have been measured with this system. The system risetime is on the order of 10ns and

is limited due to the optical detector being used.

The frequency response of the system shown in Figure 13 can be improved substantially if CS2 liquid is used. The pulse and light propagation velocity for CS2 are approximately the same. A travelling wave measurement system therefore results. The frequency response is therefore limited only by the response time of the CS2 which is on the order

of 2ps. No measurements using CS2 were taken; however, results were obtained using other liquids including nitrobenzene, water, and transformer

oil. [7]

To detector

HeNe lase> input

1.5 Pul se

lOfi H20 line ^ — Pul se

y Coaxial 1 ine conductors

Figure 13. Megavolt H

?0 Measurements (Wunsch [7 1). '2

17
5. Summary and Conclusions

Fast pulse voltage measurements are typically done using capacitive

and resistive voltage dividers. However, the time response of these systems is limited due to stray capacitance and inductance effects which

either slow the observed risetime or produce oscillations for fast pulses.

Additionally, these techniques are susceptible to ground loops and electromagnetic interference.

Electro-optical measurement techniques utilizing either the Pockels

or Kerr effects can be used to measure pulsed voltages. The Kerr effect

is more appropriate for pulsed high voltages. Kerr effect measurement

techniques have no ground loops and can be effectively snlelded from EMI.

Kerr effect measurements do not typically incorporate divider circuits.

The frequency response is therefore determined by the geometry of the Kerr

cell used. Lumped measurement systems, where the Kerr measurement cell

is capacitive, can be used to measure voltage risetimes in the nanosecond regime. Faster responses are possible using the Kerr cell is a transmission line geometry. The ultimate frequency response for this arrangement can be much less than Ins.

Kerr effect techniques, in addition to being rather noise immune

and very fast, are particularly accurate at high voltages. Because of the

quadratic relationship between the voltage being measured and the detected

light intensity, the measurement accuracy improves substantially at high

voltages.

Electro-optical measurement techniques have definite advantages for

pulsed, high voltage measurements. It should, however, be noted that they

are typically more difficult to implement than more conventional techniques. The techniques require a laser, a Kerr cell, and detector system.

Ultrafast pulses can require an image converter camera. However, for certain applications and situations, the added complications are warranted.

It is also probable that for some subnanosecond voltage measurements,

electro-optical techniques, used in conjunction with an image converter

camera, may be the only techniques with sufficient frequency response and

noise immunity which can be used successfully.

6. References

[1] R. L. Hebner, Jr.,R. A. Malewski, E. C. Cassidy, "Optical Measurements

at High Voltage Levels," Proc. IEEE, Vol. 65, pp. 1524-1548 (1977).

[2] P.D. Thatcher, "Optical Effects of Fringing Fields in Kerr Cells," IEEE

Trans. Elec. Insul . , Vol. EI -11 , pp. 40-50 (1976).

[3] E.C. Cassidy, N. N. Cones, S. R. Booker, "Development and Evaluation

of Electro-optical High Voltage Pulse Measurement Techniques," IEEE

Trans, on Instr. and Measurement, Vol. IM-19, pp. 395-402 (1970).

[4] R.E. Hebner, Jr. and M. Misakian, "Calibration of Hiqh Voltaae Pulse

Measurement Systems Based on the Kerr Effect", NBS Report NBSIR 77-1317,

Sept. 16, 1977. 1R
E.E. Bergmann and G. P. Koleogy, "Measurement of Nanosecond HV

Transients with the Kerr Effect," Rev. Sci. Instr. , Vol. 48, pp.

1641-1644 (1977).

A.J. Schwab, "Compact High Power No Laser: Circuit Theory and

Design", IEEE Journal of Quantum Electronics, Vol. OE-12, pp.

183-188 (1976).

D. C.Wunsch, A.S. Sanchez, and E.W. Green," Nanosecond High Voltage

Measurements", CPEM 74 Digest: Conf. on Precision Electromagnetic

Measurements, London, England, 1-5 July, 1974 (London, England:

IEE 1974) pp. 82-83.

E. C. Cassidy and R.E. Hebner, "Experimental Study of the Behavior

of Nitrobenzene under Varied High Voltage Conditions", Annual

Report, Conf. on Elec. Insul. and Diel. Phen:, National Academy

of Sciences, Washington, D. C, pp. 37-44 (1973).

W. Botticher, Ch. Homann , and H. Hukner, "Electro-optical Measurement of Current and Voltaqe in Fast, Hi qh Pressure Glow Discharaes

J. Phys. I: Sci. Instr., Vol. 11, pp. 248-250 (1978).

J.E. Thompson, M. Kristiansen, and M. D. Hagler, "Optical Measurement of High Electric and Magnetic Fields", IEEE Trans, on Instr.

and Meas., Vol. 25, pp. 1-7 (1976).

J.E. Thompson J. Lin, K. Mikkelson , and M. Kristiansen, "Investigation of Fast Insulator Surface Flashover in Vacuum," IEEE Trans^

actions on Plasma Science, Vol. PS-8, pp. 191-197 (1980).

T.J.Davies and M.A. Nelson, "Optical Sampling Transient Analyzer

System," presented at Conf. on Laser Enqr. and Applications,

Washington, D.C. ; 1975.

19
HIGH VOLTAGE PROBE FOR LIQUID IMMERSION

Neville W. Harris

Ion Physics Company

Burlington, MA 01803

Abstract

A capacitive voltage divider for the measurement of very

high voltage transients in liquid-filled transmission lines

has been developed. The probe is suitable for pulse lengths

in the range 10 nsec to 1 ysec, and peak voltages between

50 kV and 10 MV. The divider automatically compensates for

changes in dielectric constant and loss.

Key Words: Voltage monitor; capacity divider; pulse voltage

monitor; high voltage divider; waterline voltage monitor.

1. Introduction

The development of a very high power, short duration

pulse generator operating at voltages in excess of 400 kV

required a fast high voltage probe. The probe described is

suitable for use in liquid dielectrics at electrical field

stress levels near breakdown and compensates automatically

for changes in dielectric constant and resistivity.

The 400 kV pulse generator employed the usual technique

of charging a transmission line to twice the required voltage

and discharging into a matched load. This resulted in a flat

topped pulse with very short rise and fall times. Powers up

to 10 12 W with pulse lengths of the order of 50 nsec were

generated in this fashion. The transmission line, in this

case, was a coaxial line filled with water stressed close to

its breakdown limit. A suitable voltage monitor had to be

immune to the very severe interference fields radiated by the

generator and had not to reduce the breakdown voltage of the

coaxial line.

20
2. Principle of Operation

Figure 1 illustrates the principle of operation. A cylindrical

CIRCUMFERENTIAL

AREA A

OSCILLOSCOPE

END AREA B GROUND

HIGH VOLTAGE LINE

Figure 1. PRINCIPLE OF OPERATION

plug is mounted concentrically into a recess in the grounded

conductor of a transmission line. The plug end is flush with

the grounded conductor surface. The dielectric fluid filling

the transmission line also fills the capacitive monitor. The

plug to grounded conductor distance Is t and opposing area A.

The plug to high voltage conductor distance is d and end area

B. The voltage division ratio is then

Ad

Bt (1)

and the field division ratio is simply B/A.

If the field stress in the medium is near breakdown

level then the field in the monitor can be reduced to safe

levels by making B>>A. B should be significantly larger than

A to take care of mounting alignment errors and local field

enhancement

.

Note that the conductivity and the dielectric constant

of the fluid do not need to be- known. This means that temper

ature, impurity and frequency effects are automatically compensated. Nonlinear effects in the fluid are compensated

properly only when area B is similar to area A.

The resistor R s between the plug and the monitor cable

reduces the sensitivity of the monitor and increases it bandwidth. Care must be taken that self capacity of the resistor

or its capacity to its holder do not degrade the bandwidth.

3 Construction

Figure 2 shows the construction of the voltage divider

A cylindrical plug was mounted on an insulating sheet which

was in turn fixed to a backing plate. A resistor was

21
Figure 2. CONSTRUCTION OF MONITOR

connected between the plug and a 50 Q monitor cable. This

assembly was then inserted into a recess in the ground conductor of the transmission line and sealed with an O-ring as

shown. The diameter of the plug was close to that of the

recess, and the face flush with inner surface of the ground

conductor. Centering the plug in the recess can be accomplish

ed using grooves in the insulator as the gap t is usually In

the range 0.2-1 mm. Accurate centering may also be accomplish

ed by the use of Insulating centering screws; see Figure 3.

22
4. Electrical Characteristics

OSCILLOSCOPE

Figure 4. EQUIVALENT CIRCUIT OF DIVIDER

The equivalent circuit for the divider Is shown in

Figure 4. C]_ , Cg, C s

, and R

g are the capacity between

plug and line, capacity between plug and ground, stray

capacity between resistor R g and plug and value of the series

resistance, respectively. Cg is readily calculable from the

formula for concentric cylinders, as the end effects are very

small. Similarly with Ci, the plug and the inner conductor

of the transmission line form a parallel plate capacitor with

the ground conductor acting as a guard ring. We can assume

Cg. to be much greater than C]_. The effect of finite fluid

resistivity is modeled in the equivalent circuit by R]_ and

R

g

, note that R]_Ci = RgCg.

Neglecting losses, fluid resistivity and assuming

Cg >> C]_ the output of the divider, U, into the cable for an

input step function voltage V is

tt Ar°l 50 U = V— (R +5 0) exp

g s (R +50)C

S i (2)

The low frequency limit of the divider is determined by

the decay time constant, (R s +50)Cg. The high frequency

response of the probe is limited Dy transit time effects

along the plug and the reactance of the resistor R s

. However, if the value of R

g

is suitable chosen, (e.g., about

1 k ti) the effects of resistor shunt capacity and series inductance are negligible.

The pulse generator, for which the probe was designed,

uses water as the dielectric due to the high energy density

storage capabilities of water. However, water suffers

from the disadvantages that the dielectric constant varies

with frequency and temperature, and in particular, the

resistivity depends markedly on impurity concentration.

The divider output U for an input step function V is

more complicated than that given by Eq . (2), if we take fluid

conductivity into account.

23
U= V50_

DT+50')

1-expCYt/C^+C.

)

E

YR-,

-exp

-Yt

(C + C, ) K 1

; (3)

where Y=R,

1 + R

1 f

1 + (R

s +50) _1

For times much shorter than the decay time constant,

the divider output is equal to that given in Eq . (2). For

times comparable with this time constant, the output decays

more slowly and is not asymptotic to zero. Hence an increase

in the conductivity of the dielectric increases the bandwidth of the divider and does not affect its calibration or

division ratio.

The contribution to C

ff made by the mounting insulator

is negligible if the insulator thickness is very much

greater than the gap between the plug and the recess in

which it is mounted. Consequently, a change in the dielectric

medium affects C and C-^ in the same ratio and does not

affect the calibration. The probe blends in smoothly with

the ground conductor of a transmission line and does not

degrade the standoff voltage capabilities of the line. The

body of the probe is an .almost complete shield which keeps

out unwanted electromagnetic fields. Pickup of interference

signals is further minimized by the symmetrical, coaxial

construction of the inner conductor and series resistor.

The probe is suitable for pulse lengths between 10 nsec and

1 ysec with outputs in excess of 100 V and a signal to noise

ratio better than 35 dB.

Result s

500 kV

»

20 nsec

Figure 5. MONITOR OUTPUT

24
Figure 5 shows a typical waveform from the monitor.

When the generator was connected to a resistive load,

this capacitive monitor agreed with the current monitor

fitted and had substantially less 'hash'.

The best calibration procedure involves launching a known pulsed voltage onto the transmission line. Sensitivity

can be increased, for the purposes of calibration, by

reducing the transmission line spacing d. The alternative

method of calibration, measuring the capacities Cg and C±,

is less satisfactory due to films on the electrode surfaces.

Errors and malfunction of the monitor can be caused

by the presence of foreign matter in the body of the

monitor. In water systems, these can be either air bubbles

or solid particles. Little trouble was experienced with

either bubbles or particles as the monitor was mounted near

(but not at) the bottom of the horizontally mounted coaxial

transmission line. A crude but effective method of flushing

the monitor was simple to loosen momentarily the mounting

screws, allowing water to flow past the O-ring.

6. Conclusions

A simple capacitive voltage monitor that depends for

its division ratio on geometry, not material properties has

been described. It was used in a high power, high voltage

(400 kV) pulse generator and gave a clean reliable signal.

25
EXPERIMENTAL COMPARISON OF STEP-RESPONSE AND RAMP-RESPONSE

MEASUREMENTS IN FREESTANDING DIVIDERS

Robert E. Hebner

El ectrosy stems Division

National Bureau of Standards

Washington, D.C. 20234

It is conventionally assumed that the magnitude and the waveshape

of a high voltage pulse can be determined from measurements of the

divider ratio, the step response, and the output voltage. This

assumption has been verified experimentally by measuring the input

with a second, reference divider which introduces negligible distortion

in the applied waveform. The validity of the approach had been demonstrated for pulses ranging from less than a kilovolt to more than a megavolt on microsecond time scales. Specifically, in a system

designed to measure pulses with peak values up to 4 MV, the differences

between the calculated and measured voltages were typically less than

three percent. The testing of the reference divider, the step response

measurements, the data analysis and the high voltage testing of the

system are described.

Key words: divider; high voltage measurements; impulse; step response.

1. Introduction

Standardized techniques to calibrate impulse voltage measurement

systems are important in such applications as insuring that measurements made in a number of different laboratories are consistent or providing the buyer and the seller of pulsed voltage apparatus a technical basis for insuring that procurement specifications are met.

In recognition of these and/or related advantages, a standard

(IEEE 4) [1] has been developed to provide a consistent approach to

the high voltage measurements required by the electric power industry.

For pulse measurements, the fundamental approach used in this standard

is to determine the divider ratio--a single real number by which the

output voltage is multiplied to determine the input voltage. In addition, it is verified that the transfer characteristics are frequency independent over a frequency range suitable for the voltage

pulse which is to be measured. In general, the ratio is determined

using a low voltage technique, e.g., impedance measurement. The

frequency independence is inferred from a measurement of the step

response of the device under test. This report describes an attempt

to extend the approach to a very large system by correcting the

waveform using the measured step response of the item under test [2].

26
2. Experimental Background

The desired measurements required a variety of test and measurement circuits to be developed. The most critical component was a well-characterized divider which could be used as a reference. The

divider used was a counter-wound, resistive divider rated at 1 MV in

air. The divider ratio was determined under low voltage by measuring

the dc voltage ratio, the resistance ratio, and the step response.

The ratio was determined at high voltage by comparison with an electro-optic Kerr system.

The equivalent circuit of the divider is shown in figure 1. The

high impedance element is Rh, the low impedance element is Rl, the

terminator is Rj, R] and R2 are the resistances of the cable which

connects the divider to a measuring instrument, and Rq is a damping

resistor in the high voltage lead. To determine the voltage ratio, a direct voltage Vj was applied to the input and both the input voltage

and the output voltage V0 were measured. The direct voltage ratio

was verified by measuring the individual resistances Rq, Rh> Rl> R l

R2, and Rj. The step response was determined by applying a voltage

step to the input and measuring the response at the output terminals.

The step was generated by applying a dc voltage to the input and

shorting it to ground using a mercury-wetted reed relay.

One obvious concern with the measurement of the step response is

that the circuit in which the response is measured is not the same as the circuit in which high voltage pulse is measured. A fundamental

reason for this difference is that while the relay is typically only a few centimeters high, the item under test can be several meters high.

The different configuration leads to different capacitance and

inductance associated with the high voltage lead and thus a difference

in the response.

The ratio was determined at high voltage by measuring the same pulse with the divider under test and with a measurement system based

on the electro-optic Kerr effect [3,4,5]. The Kerr system offers the

advantages of different errors from dividers, optical coupling between

the high and the low voltage portion of the circuit, and increased

precision with increased voltage.

A summary of the ratio determinations is given in table 1.

Table 1. Summary of Standard Divider

Ratio Determinations

Measurement Method Ratio % Deviation from Average

Step Response

Kerr System

DC Voltage

Resistance

10620:1

10620:1

10590:1

10560:1

+0.2

+0.2

-0.1

-0.4

27
Fig. 1 Equivalent circuit of the reference divider including

the damping resistor Rq, the terminator Ry, and cable

resistances R-| and R2.

28
The deviations in these data are about a factor of two larger than

could be attained by very careful measurement, but are small enough for

the present application.

3. Measurements on Divider Under Test

The experimental approach is first to measure the step response of

the divider under test. Then a voltage pulse is measured using both the

divider under test and the standard divider. Using the measured output

from the reference divider and the step response of the divider under

test, the anticipated output from the divider under test is calculated.

To verify that the measured response of the divider under test is appropriate for the voltage pulse to be measured, the calculated and measured

outputs are compared.

A typical test setup used to calibrate a 4 MV divider is shown in

figure 2. Typical results are summarized in table 2.

Table 2. Typical Comparison of Calculated and Measured Waveforms

Time Calculated Measured Difference

Us ) Waveform (V) Waveform (V) % of Peak

0.2 20.7 23.7 -1 .0

0.4 61.8 63.0 -0.4

0.6 118.0 115.0 +1.0

0.8 190.0 184.0 +2.0

1 .0 271 .0 260.0 +3.0

The calculated waveform is determined by solving Duhamel's integral,

t

V (t) = X f e'U) V (t-x) dx,

o TT / i

where N is the divider ratio, V 0

is the calculated output voltage, x is an integration variable, o'(x) is the derivative of the measured

step response, and V-j is the high voltage applied to the divider.

The high voltage V-j is determined by multiplying the output voltage

from the reference divider by the reference divider ratio. For the

measured output to agree with the calculated output, it is necessary to

measure accurately the output of the reference divider, the ratio of the

reference divider, the output from the divider under test and the step

response of the divider under test. In a large number of measurements,

the disagreement between the measured and calculated output from the

divider under test ranged between 0.5% and 1% of the peak voltage. These

measurements, therefore, suggest that the divider under test is suitable

for the measurement of the waveshapes used if uncertainties of several

percent can be tolerated.

29
Fig. 2 Typical test set-up for step response and low-voltage

ramp response measurements.

30
These tests were performed using a voltage pulse which had a linear rise of about 1 us. The voltage was "chopped" at 1 ys, so that the input voltage rapidly (i.e., in a time short compared

to 1 us) fell to zero. A low voltage pulse generator, with an output impedance of 1 ft and a peak output voltage of 1 kV was used

for many of the tests. Typical measurements of a single ramp voltage

are shown in figure 3. Tests were also made at high voltage, up to

about 1 MV, by using a sphere-gap to chop a standard lightning

impulse near its peak. The inaccuracies in the high voltage measurements also ranged up to 7% of the peak voltage.

4. Summary

A technique to verify the accuracy of pulse measurement systems

for front-chopped voltage waveforms, the most difficult to measure,

according to the general guidelines of IEEE 4 has been applied to

a 4-MV system. The approach has been verified for particular sets

of conditions in interl aboratory comparisons. The technique has

been used to measure pulses longer than a few microseconds and with

peak voltages up to a few megavolts.

This work reviewed the extension of the approach to systems

designed to measure pulses with peak values up to 4 MV. Techniques

to verify accurate measurement for pulses of higher voltage and/or

shorter times have yet to be developed and evaluated. The evaluation

is important to prove that the verification technique is valid for

a range of pulse shapes, pulse amplitudes, and circuit configurations.

5. References

[1] The Institute of Electrical and Electronics Engineers, Inc.,

IEEE Standard Techniques for High-Voltage Testing , (New York:

Wil ey-Interscience, 1978).

[2] R. E. Hebner, D. L. Hillhouse, and R. A. Bullock, "Evaluation of

a Mul timegavolt Impulse Measurement System," Nat. Bur. Stand.

(U.S.), NBSIR 79-1933, 1979.

[3] E. C. Cassidy, W. E. Anderson, and S. R. Booker, "Recent Refinements

and Developments in Kerr System Electrical Measurement Techniques,"

IEEE Trans. Instrum. Meas., Vol. IM-21 , pp 505-510, 1972.

[4] R. E. Hebner, E. C. Cassidy, and J. E. Jones, "Improved Techniques

for the Measurement of High-Voltage Impulses Using the Electrooptic

Kerr Effect," IEEE Trans. Instrum. Meas., Vol. IM-24, pp 361-366,

1975.

[5] R. E. Hebner, R. A. Malewski, and E. C. Cassidy, "Optical Methods

of Electrical Measurements," Proc. IEEE, Vol. 65, pp 1524-1548, 1977.

31
Fig. 3 The low voltage measured using the divider under test

(BPA-C) , the reference divider (NBS-1) and direct

measurement wi thout an external divider.

32
6. Sponsor Acknowledgement

Partial support for this work was obtained from the Bonneville

Power Administration, U.S. Department of Energy.
A COMPARISON OF THREE DIFFERENT DESIGNS OF RESISTOR DIVIDER

T.R. McComb and M.M.C. Collins, W.J. Sarjeant*

National Research Council of Canada,

Division of Electrical Engineering,

Ottawa, Ontario, Canada, K1A 0R8

University of California,

Electronics Division,

Los Alamos National Lab.,

Los Alamos, N.M., 87545

This paper describes a comparison of dividers developed in the

Laser Physics Section at NRCC with dividers developed in the Power

Engineering Section, NRCC. Requirements on the design of high voltage

impulse measuring systems, which are to be used to measure short-duration impulse voltages, are discussed briefly using transmission line

concepts. Developments in the field of Power Engineering are summarized

and the classification of impulse measuring systems into two-component

and three-component systems is outlined. Four different dividers, whose

response times lie in the range 1.4 - 15 ns, are described. Comparative

measurements of the crest value of linearly-rising impulses chopped on the front have been made at a single voltage; these measurements have

been corrected for the effects of system response time. Measurements of

the crest value of a trapezoidal pulse have been made using three of

these dividers and the results are presented in the form of a calibration

curve covering the range 100 - 400 kV. Estimates of the correction

needed to allow for the dividers' response times are calculated on the

basis of rectilinear approximations to the trapezoidal pulse.

Key Words: Impulse measuring systems; resistor dividers; comparative

measurements; design; dividers; voltage measurement; response time errors

.

Research over the past 30 years has led to the development of new International and American Standards [1,2] on high voltage test techniques. The recommendations of these standards for impulse measuring

1 . Introduction

* Formerly with National Research Council of Canada, Division of Physics,

Ottawa, Ontario, Canada, K1A 0R6

34
systems are based on the measurement of the Standard Lightning Impulse

whose waveform is given by

v = k(e

-at

(1)

where a = 1/68.5 ys and 3 = 1/0.405 ys and k is the efficiency of the

generator.

This gives a virtual front time (1.67 times the time interval

between the instants when the impulse is 30% and 90% of crest value) of

1.2 ys and a time to half- value of 50 ys. The accuracies of measure- ment required by these standards are crest voltage ± 3%, front time

± 10% and time to half- value ±10%. Accepted procedures for, and advice

on, meeting these requirements are given in the standards.

The purpose of an impulse measuring system is to provide a reduced

copy of the input voltage waveform and this requires that the ratio of

the system is constant throughout the frequency range of interest. The

design of a satisfactory measuring system entails determining this

frequency range and ensuring that the ratio of the system is constant

for this range. In practical circuits high voltage impulses are generated using switching gaps and these gaps generate fast pulses which

propagate through the circuit. Whether oscillations are generated and

how long they persist is mainly determined by the reflection coefficients

in the system. A full analysis of the circuit involves treating interconnections as transmission lines and treating apparatus as transmission

lines for twice the travel time of the apparatus [3,4,5].

An impulse measuring system can be thought of as a two component

system - a transmission line (the high voltage lead) and a termination

(the divider, recorder, and the cable interconnecting them). The

divider impedance will not usually be equal to the line impedance and

hence there will be a reflection generated at the divider which will

lead to oscillations in the circuit. These can be reduced by the insertion of a terminating resistor in the high voltage lead at the test

object and this gives a three component system. When the terminating

resistor matches the surge impedance of the lead no oscillations will

be generated in the measuring system.

Because of the response time of the measuring system the output

will not immediately follow the input. There is no general method of

applying a simple correction for this. In the special case of a linearly rising voltage it can be shown that the error in amplitude 6(t) at

time (t) is given by

2. Errors Caused by the Response Time of the System

6(t) = S /* [1 - g(t)] dt

35

(2)
[Appendix 4B of reference 2] where S is the rate of change of voltage

and g(t) is the normalized step response. After a sufficient time, known

as the settling time, 6(t) is approximately equal to 6(°°). For any

linearly rising voltage which lasts longer than the settling time the

value of S can be read from the output and the necessary correction

calculated.

Standards impose further restrictions on the response characteristics when the system is used to measure impulses chopped on the front.

The initial distortion time (T0

- illustrated in Figure 1) must be less

than one twentieth of the time to chopping.

o VIRTUAL ZERO

FIGURE I UNIT STEP RESPONSE - DEFINITION OF TERMS

T0 - INITIAL DISTORTION TIME TQ - PARTIAL RESPONSE TIME

Tn

= Ta - T/3 + T

y - T 8 . . . EXPERIMENTAL RESPONSE TIME

3. Design of Resistor Dividers

3.1 Principles of Design

An ideal resistor divider would have a high voltage arm and a low

voltage arm whose impedances are purely resistive in the frequency range

of interest.

In practice this condition cannot be achieved for the high voltage

arm because the resistor will have distributed stray capacitance to

ground. For a resistor mounted vertically on the ground plane the distribution of capacitance is non- linear and the stray capacitance of the

lower parts of the high voltage arm will have to charge through the

upper parts of the resistor giving a long settling time. The addition

of a shielding ring or plate at the top of the high voltage arm will

provide a capacitative path for this charging current. The shape and

36
position of the shield (or shields) can be adjusted to give a more uniform field distribution which not only improves the response characteristics but also increases the voltage rating [6].

A purely resistive low voltage arm is readily achieved by using a good quality, high frequency, coaxial cable terminated in a resistor

whose resistance is equal to the surge impedance of the cable. When

direct input to the deflection plates of the oscilloscope is used, as in

this work, a damping circuit must be inserted between the oscilloscope

plates and the cable terminating resistor.

The total resistance of the divider must be sufficiently high to

avoid excessive loading and to provide a suitable division ratio. However, the distribution of stray capacitance of a well designed screened

divider will still have some non-linearity and this residual effect will cause the divider response time to increase with divider resistance. A

nominal value of 25 ft/kV [6] offers a practical compromise on the con- flicting requirements of acceptable loading and fast response time.

3.2* Resistor Dividers

These principles have been applied to the design of four dividers

which are shown in Figure 2. The low voltage arm for each of these

dividers was 30 m of RG17/U .terminated in an 8 W 52 ft carbon composition

res i s tor.

A

i

Mr*" -

(a) (b) (c) (d)

FIGURE 2. RESISTOR DIVIDERS

(a) WIRE - WOUND, AIR - INSULATED DIVIDER

(b) WIRE - WOUND, PRESSURIZED DIVIDER

(c) SILICON CARBIDE DIVIDER

(d) COPPER SULPHATE DIVIDER WITH MINI-PROBE

37
3.2.1 Power Engineering Section Dividers

The wire-wound, air-insulated divider shown in Figure 2(a) was designed to measure full lightning impulses and lightning impulses chopped

near crest [7]. The high voltage arm was a wire-wound resistor with a non-inductive winding. Conical shields were mounted at the top and at

the bottom of the high voltage arm. The divider is rated at 350 kV for

standard lightning impulses.

The pressurized divider shown in Figure 2(b) was designed to

measure chopped impulses with risetimes of from 50 ns to 1 ys [7]. This

divider is a faster version of the ai r- insula ted divider and was made

by reducing the divider size by a factor of eight and insulating it in

is an atmosphere of nitrogen at a pressure of 300 kPa [7]. The

pressurized divider is rated at 350 kV for impulses chopped at 1 ys.

3.2.2 Laser Physics Section Dividers

Two different dividers were developed for. use in pulsed laser

systems to measure impulses which are typically trapezoidal in shape

and have durations from 100 ns to 5 ys. One of these used a silicon

carbide resistor as the high voltage arm, shown in Figure 2(c), while

the other used a copper sulphate resistor, shown in Figure 2(d).

The silicon carbide divider used a Union Carbide Type AS 6.8 kft resistor which was impregnated with, and contained in, a silicon

dielectric fluid (Dow Corning DC-200, viscosity 20 mm

2 /s) to reduce

local damage at high voltage stress. The resistor length was 0.45 m and

the value after impregnation was 10 kft. Bulk silicon carbide resistors

offer high power dissipation and low inductance but they can be non- linear at high stress levels. This divider was used with a flat shield

and a damping lead which consisted of six Allan Bradley type HB 2W

carbide composition resistors each 10 ft and each separated by 6 in. of

#22 AWG wire and encased in Tygon tubing which had been filled with

silicon oil. This divider was rated at 300 kV for pulses shorter than

5 ys.

The copper sulphate divider was a two-stage divider with the second

stage using the terminated cable as a low voltage arm. The high voltage arm of this divider consisted of a solution of copper sulphate in

distilled water contained in a length (0.3 m) of Tygon tubing (i in.

I.D.). Connections were made by immersing cylindrical rolls of high

purity copper wire gauze in the solution and the solution concentration

was adjusted to give a resistance of 3.0 kft ± 5%. The low voltage arm

was a mini-probe [8] in parallel with one or more 330 ft carbon 2W re- sistors as required to limit the voltage on the mini -probe to less than

20 kV. The high voltage arm of the mini -probe consisted of a lossy

transmission line terminated in a carbon 2W resistor. This resistor was partly covered by a grounded shield and the shielding was adjusted to

give the best response time (estimated as 30 ps [8]). When immersed in

oil, this two-stage divider was rated at 400 kV for pulses shorter than

1 ys

.

38
4. Impulse Measuring Systems

4.1 Common Components

The cables, matching circuits, recording units, and ground returns

were the same for all systems used. Each cable was 30 m of RG17/U

terminated in a low-inductance 'cartwheel', made from four 220 Q, (10%)

carbon 2W resistors selected to give a 52 Q, total resistance. Experience indicates that these resistors must be pulse conditioned before

use. Oscillations in the connections between the terminating resistance

and the oscilloscope deflection plates are damped by the circuit. The

surge oscilloscopes were Southern Instrument (circa 1960) oscilloscopes

which had been extensively modified at NRCC to improve their shielding

and linearity [9].

4.2 Matching Resistor

The ordinary type of non-inductive resistor, obtained by winding

two resistors on the same former in opposite directions, was unsatisfactory when used as a matching resistor in a three component system

because of its residual inductance. A satisfactory design [3] was achieved by the use of four such resistors in parallel, arranged in the

form of the spokes in a wheel, the hub forming one terminal and the rim

the other. To reduce the stress on each resistor, two of these cartwheels were used in series.

4.3 Grounding

The entire circuit was mounted on a large ground plane (7.8 m by

13.5 m) which consisted of interconnected rectangular sheets (1.3 m by

2.7 m) of brass or aluminum. These sheets could be easily moved and

clipped together to give a ground plane appropriate to the circuit in use. Each oscilloscope was grounded directly by the sheath of the RG

17/U to the ground plane at the base of the divider being used with it

and indirectly by the stray capacitance to ground. Each oscilloscope

was isolated from the line by inserting a choke in each power lead and

by an isolation transformer. No information is available on the nature

of the danger posed by fast pulses travelling through ground systems

although this is a common problem in high voltage laboratories and gasinsulated sub-stations. In this work the operator was either isolated

from other grounds or from the oscilloscope during the impulse discharge

ti me

.

4.4 Description of Systems

The main features of each system are given in Table 1. The wirewound, air-insulated divider and the pressurized divider were used in

three component systems with a high voltage lead of bare wire (#14 AWG).

Both the laser physics dividers were designed to be used as voltage

probes without any additional leads and they were used in this manner

for some of the measurements. For the calibration with trapezoidal

39
impulses they were used with an additional wire lead to ensure that

there were no proximity effects.

TABLE 1 IMPULSE MEASURING SYSTEMS

System

Identi f i cation

II (a)

(b)

III

IV

V

VI

VII

Di vi der

Used

Wi re-wound,

Insulated in high

pressure Nitrogen

Wi re-Wound

Ai r-Insulated

Wi re-Wound

Air-Insulated

Silicon carbide

Silicon Carbide

Copper Sulphate

and secondary

divider

Copper Sulphate

and secondary

divider

No. of

Components

Terminating

Resi stor

(n)

215

215

235

432

Length of

H.V. Lead

(m)

2.1

2.1

1.8

No additional Lead

2.4

No additional Lead

2.3

5. Measurements

5.1 Response Time Measurements

A voltage step with a risetime of less than 1 ns was applied to

each impulse measuring system using a mercury-wetted reed relay as prescribed in the standards [1,2]. The response was recorded using a Tektronix WP2200 Transient Digitizer System which converted the analog

signal into 9 bit words with 512 words per sweep at sweep speeds down to

5 ns. This system was used to normalize the responses and integrate the

unit step response to give the response parameters listed in Table 2.

Corrections for the travel times of the additional vertical lead were

made using standard procedures [1,2]. Measurements were also made using

a high voltage step generated with a high pressure chopping gap [7].

5.2 Ramp Measurements

A linearly rising pulse was generated using the circuit shown in

Figure 3. This circuit was laid out in an approximate straight line and

the reference measuring system was connected in the same line. The

other measuring systems were connected, one at a time, at right angles to the reference system to minimize coupling betweer the measuring systems.

40
TABLE II RESPONSE CHARACTERISTICS

System

Response Time

(/

0

Ts P-g(t)]dt)

Settl ing

(V

ns ns

T

X - c . u DO

II (a)

(b)

7.6

7.8 85

III 10 40

IV 4.5 40

V -3.5 200

VI 1 .5 15

VII -6.3 280

* equivalent partial response time [21 is 0.3 ns

FIGURE 3. CIRCUIT TO GENERATE

LINEARLY RISING IMPULSES CHOPPED ON THE FRONT

A = 3.8 m B = 2.9 m C = 1.4 m Dond E HIGH VOLTAGE LEADS

Rise Time

(T

1 0%" T

9 0%^

ns

1.7

12

28

19*

3

1.7

2.5

Ramp measurements were performed for linearly rising pulses with

crest values of ± 100 kV and response time corrections were made.

Illustrative records are shown in Figure 4 and results are given in

Table III.

5.3 Calibration With Trapezoidal Impulses

For a comparative calibration to be of value, it must be carried

out with the impulse shapes with which the divider being calibrated is

to be used. The usual working pulse for the laser physics dividers is

trapezoidal in shape, with extremely fast risetimes and fall times,

achieved by using oil-immersed gaps for chopping. Similar pulses are generated by the circuit shown in Figure 5 but the risetimes and falltimes of the pulse are slower than those of the working pulse because of

41
the increased breakdown time of the air gaps. As the crest voltage was increased the trapezoidal pulse became more triangular in form and this

is illustrated in Figure 6. Comparative values are plotted in Figure 7, By using rectilinear approximations to the impulse the approximate size

of response time errors can be calculated and these corrections are indicated in Figure 7.

TABLE III RAMP MEASUREMENTS - CORRECTED MEASUREMENTS

Reference

System

II (a)

I

I

I

System being

Cal i brated

II

IIIV

VI

(b)

(b)

Deviation from

Reference System (%)

0.8

-1.9

2.7

3.6

Standard

Deviation {%)

0.5

2.0

0.3

0.5

s

(a)

113 kv

I05kv

96 kv

117 kv

IIO kv

103 kv

112 kv

104 kv

95 kv

(c) (d)

FIGURE 4. RAMP MEASUREMENTS

(a) WIRE-WOUND, AIR INSULATED DIVIDER

(b) WIRE-WOUND, PRESSURIZED DIVIDER

(c) SILICON CARBIDE DIVIDER

(d) COPPER SULPHATE DIVIDER WITH MINI-PROBE

42
SIX STAGES

OF

IMPULSE

GENERATOR

I80fl

24O0A

700pFi

550 a — 0

o

75 cm SPHERE GAP

2 5 cm SPHERE GAP Q

FIGURE 5. CIRCUIT TO GENERATE TRAPEZOIDAL IMPULSES

j 490 kv

410 kv

326 kv

245 kv

(a) (b)

FIGURE 6 CALIBRATION IMPULSES

(a) TRAPEZOIDAL (b) TRIANGULAR

6. Calibration and Accuracy

Each oscillogram is calibrated by the addition of a baseline and

one or more voltage lines supplied from a dc calibration source whose

total error is < 0.03%. The non-linearity of the screen is < 1% for

deflections of < ± 300 V. The effect of this non-linearity can be

further reduced by using two calibration lines which straddle the crest

val ue.

The oscillograms were read on an optical comparator at a magnification of 10. The resolution of the optical comparator was 10 _lt inches

and the accuracy was approximately 2 • 10 _Lt in. Reproducibility of a single setting was 4 x 10" 14 in. This led to approximately 0.1% maximum

reading error for peaks which caused a deflection of approximately half

43
400

300

UJ

I—CO

>-CO

200

100

MEASURED VALUES

CORRECTED VALUES

I I 100 200 300

SYSTEM II! (kv)

(a)

400

300

W 200

t—CO

>-CO

100

400

MEASURED VALUES

CORRECTED VALUES

J I I

FIGURE 7. CALIBRATION CURVES

(a) SILICON CARBIDE DIVIDER

(b) COPPER SULPHATE DIVIDER

WITH MINI-PROBE

100 2Q0 300

SYSTEM III (kv)

(b)

400

full scale. System error has been evaluated by comparing measurements

made by two versions of System II. The measurements by the quasiidentical systems disagreed by an average value of 0.8% with a standard

deviation of 0.5%. This error has been shown to be caused by the

oscilloscopes. The standard deviation is caused mainly by the ripple on the high voltage supply to the CRO tube but no explanation has been

found for the average discrepancy being non-zero. This error is small

compared with the errors caused by the response time of the calibration

with the trapezoidal pulse. For the corrected values obtained from the

ramp measurement this difference in the oscilloscope is significant.

Each oscillogram was calibrated with approximately 30 cycles of a stable oscillation at a suitable frequency. This reduced the error in

the time scale to approximately li% of the full scale deflection. The

total error in the response times is estimated at approximately 5%.

7. Conclusions

These measurements show that, for the case of linearly rising

impulses, whose duration is longer than the settling time of the measuring system, corrections can be made for errors caused by the response

time of the measuring system. These corrections give results which

agree to within 2 to 4%. However for trapezoidal and triangular impulses with oscillations no simple correction can be made and differences

as high as 15% at 400 kV have been measured. Approximate calculation

of errors caused by the response time give values as high as 7% but these

can only be taken as indicative of the order of magnitude of the

corrections needed.

44
References

[1] International Electrotechnical Commission - Standard 60 - High

Voltage Test Techniques.

[2] IEEE Standard Techniques for High Voltage Testing. IEEE Standard

4 - 1978.

[3] F.C. Creed, T. Kawamura and G. Newi , Step Response of Measuring

Systems for High Impulse Voltages. IEEE Transactions on Power

Apparatus and Systems, PAS-86, No. 11, pp. 1408-1420, Nov. 1967.

[4] F.C. Creed and M.M.C. Collins, The Systems Concept in Generating

High-Voltage Impulses. IEEE Transactions on Power Apparatus and

Systems, PAS-89, No. 7, pp. 1383-1387, Sept/Oct. 1970.

[5] F.C. Creed and M.M.C. Collins, Transient Impedance of High-Voltage

Impulse Generating Systems. IEEE Transactions on Power Apparatus

and Systems, PAS-89, No. 7, pp. 1387-1393, Sept/Oct. 1970.

[6] R.F. Goosens and P.G. Provoost, Sources of Error in the Recording of

High Surge Voltages by Means of the Cathode-Ray Oscillograph. Bull.

Schweiz: Elektrotechn . Verein, 37(7), pp. 175-184, 1946. Translated in NRC Report TT-552.

[7] F.C. Creed and M.M.C. Collins, The Measurement of Short-Duration

Impulse Voltages. IEEE Transactions on Communications and

Electronics, CE69, pp. 621-630, 1963.

[8] W.J. Sarjeant and A.J. Alcock, High-Voltage Probe System with Subnanosecond Rise Time. Rev. Sci . Instrum. Vol. 47, No. 10, pp. 1283-

1287, October 1976.

[9] F.C. Creed, An Improved Cathode-Ray Oscillograph for Impulse Recording. Bulletin of the Radio and Electrical Engineering Division,

National Research Council of Canada, Vol. 11, pp. 7-11, Oct/Dec.

1961

.

45
A 100-kV, 2-ns RISETIME, dc-COUPLED PROBE

J. Power, W. Nunnally, and D. Young

Los Alamos National Laboratory

Los Alamos, New Mexico 87545

ABSTRACT

A high-voltage probe has been designed and constructed; it is unique

because it may be used for both high-voltage dc and for transient meas- urements. The probe is rated at 100-kV dc with a short transient rating

of 150-kV peak. System risetime is less than 2 ns. The input impedance

is 1000 Mn shunted by 13 pF. A distributed RC network providing the

desired attenuation is housed in a 25.4-cm-diam cylindrical housing. A

15.2-m length of coaxial cable connects the probe to a termination network at the oscilloscope input. The oscilloscope input impedance must be

at least 100 kn to maintain the correct dc attenuation ratio.

Key words: dc-coupled probe; 100-kV rating; 2-ns risetime

1. Introduction

A high-voltage probe was needed to study the electrical breakdown

characteristics of various gases. The measurements to be made required a 100-kV probe with a risetime of about 2 ns and with dc response. Therefore, an effort to develop such a probe was initiated.

Most high-voltage probes are either resistive or capacitive dividers, as illustrated in Figure 1. Both of these type of probes can be

built to have good risetimes, but neither is appropriate for both dc and

fast transient measurements. For good transient response, the resistive

divider must be made to have a resistance value that is far too low for

high-voltage dc measurements. Purely resistive and capacitive probes are both simple types of a more general divider in which the voltage division

is accomplished by any arbitrary network such that the division ratio is

constant as a function of frequency. A new probe was developed based on the simple schematic in Figure 2. There are essentially two probes in

parallel, a high-resistance dc probe and a much lower impedance, RCcoupled ac probe. The probe is most easily understood by considering it as made of three similar sections. First is the high-voltage section

46
KZ(S)

2(S)

Figure 1. Resistive, capacitive, and general dividers

> 50 MSI

80 pF

Figure 2. Simplified probe schematic

47
consisting of the 1000-Mfi resistor in parallel with the 1000-a resistor

and the 4-pF capacitor. Section two is a coupling network including the

coaxial output cable. Section three is the termination and the output is

viewed across this section with a high input impedance oscilloscope. The

dc portion, consisting of the high-resistance components, functions as a standard resistive divider. The ac probe is made up of three RC networks

in series, each having the same time constant, and hence, the division

ratio is constant as a function of frequency. There are no small resistors and capacitors rated to 100 kV, so the high-voltage section is

really 10 identical sections in series, as shown in Figure 3. The highvoltage section is housed in a 48.3-cm-long, 25.4-cm-diam aluminum cylinder and must be viewed as a distribution network for proper high-frequency analysis.

INPUT

5CQ OUTPUT

Figure 3. High-voltage probe divider network.

48
2. Computer Analysis

The probe was simulated using a NET-2 circuit analysis program.

Figure 4 shows the model used for each of the 10 high-voltage sections,

as well as the input and output sections. The model also included a 15.2-m-long, 50-fi transmission line, which is not shown in Figure 4.

Pertinent stray capacities and inductances are included in the analysis.

The component values of the ac portion of the probe were optimized based

on the computer prediction response. Figure 5 shows the predicted output

using a 300-ps risetime, Gaussian input pulse. The probe is designed to

drive a high input impedance oscilloscope plug-in with a limited bandwidth; hence, the initial ringing at about 1200 MHz is not observed.

Based on the computer output, one would expect a minimum risetime of

about 1 ns.

3. Prototype Testing

A prototype probe was built to test the computer predictions.

3.1 Probe Components

Standard components are used in the high-voltage divider. The

capacitors are 40-pF, 5-kV, NPO ceramic rf capacitors, which were tested

100 17 40 pF rrrm.

Q

rrrm

HIGH FREQUENCY

MODEL FOR INDIVIDUAL

PROBE DIVIDER SECTION

PULSE GENERATOR AND

INPUT COUPLING 6

10 PROBE SECTIONS IN CASCADE

jrrm_ -VW\r

> OUTPUT

Figure 4. Model used for probe high-frequency analysis.

49
0 2 4 6 8 10

TIME (1 ns/Div)

Figure 5. Computer model output.

to 20-kV dc. The resistors are 100-12, 1-W Allen-Bradley carbon composition units. These were tested to 30 kV in a narrow pulse RC discharging

circuit. All high-voltage divider components are housed in an oil-filled

acrylic tube.

3.2 Pulse Generation and Input Coupling

Fast risetime pulses were generated using a mercury pulser with two

Tektronix 113 delay cables used as the charge lines. This apparatus

gives sub'nanosecond risetime, 1500 V pulses into 50 n .

Coupling to the probe was made through a 25-ft transmission line

tapered to the 25.4-cm-diam of the probe body. The transmission line was terminated with 25 ft instead of 50 ft to lower the source impedance

driving the probe. A resistive divider was built into one of the terminating resistors of the transition section. This was done to monitor the

actual input to the high-voltage probe. The risetime of the monitor

probe is about 300 ps, as shown in Figure 6.

3.3 Output Cable Termination

The probe output cable is terminated with a 50-ft series resistor, a 0.1-ft resistor, and a 40-nF capacitor in series to ground. The output is

viewed across the 0.1-ft resistor and the 40-nF capacitor. The oscilloscope input is shunted with a 100-kft resistor. All attempts to construct

the termination from standard components, i.e., 1/4-W resistors and leadless ceramic capacitors, have proved fruitless because of resonances in

50
0 1 2 3 4 5

TIME (0.5 ns/Div)

Figure 6. Transition cone monitor output response.

the 5- to 30-MHz range. The termination was then fabricated on a ceramic

disk by the Los Alamos National Laboratory Hybrid Facility. The termination is made of 10 parallel arms radially soldered to a small circular

ceramic substrate. The individual resistors are 1.0 ft each and the

capacitors are 4 nF each. No termination resonances below 1 GHz have

been observed with the hybrid termination.

Having solved the ringing due to the termination, it was discovered

that the probe input circuit was ringing because of the input capacity

and the coupling inductance. Ferrite beads were put in series with the

high-voltage section to damp the resonance, but this severely limited the

risetime, as shown in Figure 7. It was finally necessary to add 150 q of

series resistance to the input of the probe to damp this resonance. Figure 8 shows a superposition of the input and output waveforms of the

high-voltage probe. The risetime is about 1.5 ns, which is the risetime

of the Tektronix 7A16A vertical plug-in. attempts to use a 900-MHz FET

probe were unsuccessful because of severe ringing in the 100- to 300-MHz

range. The difficulty lies in coupling the hybrid to the probe input.

Some additional circuitry is required between the hybrid termination

and the oscilloscope input. This circuit serves to flatten the first

100 ns of the output pulse and to flatten the 1- to 50-ms region in which

the response switches from the ac probe to the dc probe. This circuit is

shown in Figure 9.

When properly adjusted, the response is flat within 2%, referenced

to the dc level. The attenuation ratio is about 12,000 to 1. The com- pensation network must be adjusted for a particular plug-in for optimally

51
4 8 12 16 20

TIME (2 ns/Div)

Figure 7. Probe response with ferrite damping.

INPUT

0 2 4 6 8 10

TIME (1 ns/Div)

Figure 8. Probe input and output response.

52
— PROBE 510 500 K OSCILLOSCOPE

Figure 9. High-voltage probe termination and oscilloscope

coupling network.

flat response. The overall accuracy is also limited by the oscilloscope

and the plug-in combination, which have a typical accuracy of 4%.

3.4 high-Voltage Testing

The probe was tested to 105-kV dc for 4 h to check for arcing and

heating problemSo The probe body was pressurized with 30 psig of nitrogen for this test. The input to the probe was covered with oil to prevent external insulator arcing, and no arcing was observed throughout the

test. The temperature rise of the high-voltage divider was approximately

20°C. This results in a dc response change of 3 to 4%.

A high-voltage pulse was generated with a spark-gap-switched cable

pulser that generated 60- to 70-kV pulses. The pulse amplitude could not

be accurately controlled in this pulser. The pulse risetime was limited

to 3 ns because of the inductance of the spark gap. In observing the

output pulse with the high-voltage probe, no frequency response changes were noted for the 20-ns pulse length.

4. Conclusion

A high-voltage probe has been developed for both dc and high-frequency voltage measurements at the 100-kV level. When properly compensated, the probe risetime is less than 2 ns. The probe must be connected

to low-impedance, low-inductance systems for proper operation. For

extended operation at high dc voltages, it is necessary to cool the oil

insulating the high-voltage divider to eliminate temperature stability

effects, which are about 0.16%/°C. The accuracy of the probe alone is

approximately 2% neglecting temperature drift and amplifier accuracy.

53
Discussion Session 1

Richard Malewski (IREQ). The first comment is associated with the

presentation of Mr. Harris who has shown design of a voltage probe

used in a coaxial line with liquid dielectric. We had a similar

problem in measuring fast transients in a coaxial bus filled with

SF5. There are some common difficulties in these two applications.

The schematic circuitry is shown in figure la. There is a high

voltage bus and a probe which possesses some capacitance to the high

voltage bus. This probe in the design of Mr. Harris' has a capacitance

to the grounded envelope. If we look at the probe, we realize that

there is a current flowing to ground through the capacitors which

produces a magnetic flux. In a physical layout of this kind, the

inductance of the low voltage section of the circuit is fairly high

in comparison to that of the high voltage part with no damping

whatsoever. So an equivalent schematic is like that shown in

figure lb with an input high voltage, a capacitor, an integrating

network and the coaxial cable. But, in the first loop, we have

stray inductance and no damping if R] , Rg = 0. So obviously, the

current in this loop will be of an oscillatory character with the

frequency determined by capacitance C"| , Cg and inductance L] , Lg.

Mr. Harris proposes to simply smooth the" response by inserting an RC circuit with a time constant which will smooth the oscillations.

That's a fairly practical way out of it. A different approach is to

fashion the low side capacitor from a plastic cylinder which has

been painted with resistive paint. Now, the current which originally

flowed through a straight inductance will have to circulate in this

resistive layer. The damping will make this current aperiodic and

eliminate the need for the RC element. Another approach was proposed

by people from the University of Munich. They did not use the paint,

but they put ferrite cores in the circuit introducing loss at high

frequency. I believe it is better to control the current flowing

through the divider and then to see the actual voltage drop across

the low voltage arm.

I have another comment regarding the experiment described by

Hebner in which he compared two dividers, one a large high voltage

divider and the other one a small precise reference divider. A

similar exercise in our lab used a bigger divider which was 20 m in

length. The length of the lead to the small divider was 25 m. When

we compared the outputs from the two dividers, we obtained different

results for different applied pulses, i.e., a square pulse or a front

chopped triangular pulse. We realized that with pulses of duration

of a few hundred nanoseconds, the frequency spectrum extends to fairly

high frequencies and we are in a range where the connecting lead may

be a quarter wavelength of some of the harmonic frequencies of these

pulses. The lead is then long enough to be considered as an effective

transmitting antenna with frequency selective features which can effectively radiate energy at a given wavelength. A simple calculation

indicates that some pulses we are recording have a frequency spectrum

54
HV CONDUCTOR

(b) HV CONDUCTOR

"A"

Rs

Rg:

i Lg

Cs4=

To

Scope

Figure 1. (a) Physical configuration of capacitive di

(b) Lumped parameter equivalent circuit.

55
rich enough to add significant contribution at the harmonic corresponding

to this 1/4 wave transmission from the lead. The only approach we tried

to eliminate this problem was to replace the copper wire by a resistive

wire and vary the distribution of the resistance along the lead.

This improved the performance a bit, but I should admit that at this

stage we do not have a ready made answer for this problem. Perhaps

the third comment is about what Dr. McComb said about the dividers

used in their Physics Department. They are using some exotic materials

such as liquids or carborundum-type resistors for the high voltage

part of the device. A divider is a measuring device which is supposedly

linear, both in voltage and in frequency. The dielectric constant of

a liquid of the kind used for a divider varies with frequency. We can represent a divider which has a high voltage part in the form of

a column filled with liquid as a parallel connection of resistance

and capacitance. The capacitance varies with frequency and since the

liquid has in general a high dielectric constant it is not negligible.

In a resistor divider, the ground capacitance is usually predominant.

This is not true in the case of a liquid type of divider because of the

large capacitance of the high voltage element. Therefore, the

variations of this capacitor with frequency results unvariably in a distorted response. Even a worse situation occurs .when we use the

type of resistors which are basically developed for lightning arrestors.

These are very non-linear resistors and the response of such a resistor

always indicates some small irregularities. Why? Because these

resistors are composed of granules of carborundum enveloped in thin

layers of silicone insulation and the conduction in such a divider

is based on breakdown of these small layers of insulation which

connects the conductive carborundum particles. These materials are non-linear and will always exhibit small variations of resistance as

a function of time. So, to me, it is not a very promising idea to use the thyrite resistors or liquids for high voltage arms.

Neyj 1 1 e Harri

s

(Ion Physics). I am at a slight disadvantage since I have only made my probes for water lines. I would comment that

Mr. Malewski has got the right diagram. However, the equivalent

circuit is not right. He talks about a large inductance Lg and a small one L]. I think you will find this the other way around. The

low voltage capacitor is rather like a coaxial line with a very low

impedance and an inductance which is very very small. The inductance

coming from the main gap is much larger. So in the divider circuit,

the inductance L] is much larger than the inductance Lg. I agree

that they form resonant circuits. However, I did not have any capacity

at point A (see figure lb). There is stray capacity, but there was

no integrating capacitor. The cable leading to the oscilloscope was matched and hence appears as a pure resistance. Where you get most

of the trouble is in the resistor Rs. The inductance and the stray

capacities of this resistor do, in fact, give most of the trouble. I did not try various materials. I used stainless steel which happens

to be a fairly resistive material. The water losses, Rg, also serve

to damp resonances. My opinion- is that any oscillations you get are more associated with the series resistor Rs.

56
Malewski . Could you indicate the current flow through the divider,

how the current flows from the high voltage bus to the ground through

the divider?

Harris . Most of it flows through the capacitors Ci and Cg as shown

in 1(a). There is damping due to the water line since water has a loss tangent and also conductivity. As you increase the capacity Cg,

then for the same bandwidth (50 + R s )Cg, you need only a low value

for the resistor R s. The impedance of this oscillating circuit is

/Lg/Cg which should be compared with the damping resistance,

(R s + 50). Thus, if you make Cg large the ratio of damping resistor

to the circuit impedance increases as /Cg. Thus in practice, the

circuit is damped by a parallel resistance rather than a series

resistance as proposed by Mr. Malewski.

Bob Hebner (NBS). The second comment that Richard had to make was

about my paper, and I want to thank him very much because what he has

done is reinforce the point that I was trying to make. I had mentioned

that we know the technique worked well for voltage of the order of a megavolt and for pulses longer than microseconds. We have some serious concerns that it may not work well outside of those regimes.

I have shown how, under a specific condition, we extended that regime

slightly. Richard has given a concrete example of how, when they

attempted to extend that regime, the technique did not work. And it

shows that there is definitely a need for a better measurement

technique and it shows that it is naive to assume that the

techniques that work in one range of frequencies and voltages will

apply to all ranges.

Terry McComb (NRC, Canada). I think that a major point in regards to

both the si 1 i cone carbide and the copper sulfate is that they are extremely cheap and rugged devices. We recognize that they are nothing like as good as wire-wound resistors as we showed in our results. However, they can be set up quickly, and we can overvolt

them when we are playing in the lab. But, as Richard mentioned, they

are certainly not ideal. We have a long way to go to extend and

develop pulse measuring probes for very fast pulses.

Jim Sarjeant (Los Alamos). Richard, I would like to respond to your

comments in a general way for a moment if I may. My impression here,

listening to the talks given this morning, is that we are discussing

two different categories or classes of voltage measurement needs.

One is the extremely high accuracy calibration type of requirement

such as the work that is going on at NRC, NBS, and various other

places around the world and quite effectively over the last 20 or

25 years. The second class of application is where you have a device

in the laboratory and you would like to measure its electrical

parameters to some reasonable degree of accuracy and reproducibility.

And, indeed, I feel that the bringing together of these two categories

of diagnostics is one of the key points of this particular meeting.

57
Copper sulfate dividers for very fast voltaqe measurements have been

around now for nearly 50 or 60 years. Units have been built by people

at Physics International with response times of under 100 picoseconds.

They have been used at voltage levels of the order of 100 kilovolts

per inch. The response times when using dc type or long pulse

durations pulses were measured and none of the behavior that Richard

has indicated was observed. I would like to point out that when we made our measurements at extremely high stress levels in distributed

systems we did indeed see the behavior that Richard referred to. So,

in conclusion, we are talking about two classes of operation. One is

something that is big and tall and the other is something that is

short and small. In the laboratory, you may have a laser system

operating as high as a megavolt. You would like to have a compact

voltage monitoring device whose intrinsic response time and resonance

in that system are much faster than any of the characteristic

frequencies of your system. It was for such applications that Terry

and I spent some time comparing the real devices that people have

been using for the last 10 to 12 years to the highly accurate dividers

used in the high voltage field.

58
CALIBRATION OF CAPACITIVE VOLTAGE PROBES

IN WATER-DIELECTRIC, HIGH POWER PULSE GENERATORS*

M. Wilkinson, E. Chu

Maxwell Laboratories, Inc.

8835 Balboa Avenue

San Diego, CA 92123

Capacitive voltage probes are frequently used in the water pulselines of high power pulse generators. These usually consist of small

electrodes isolated from the outer conductor of the pulseline. The

capacitance between the electrode and the inner conductor determines

the output of the probe.

Due to finite resistivity of the water dielectric, a fast, high

voltage pulser is usually required to calibrate these probes. We will

discuss a method, which we call the "capacitance-current" method, that

allows probe calibrations with a slow pulser independent of the resistivity of water. We will compare calibrations obtained using this

method against those obtained using more conventional methods.

*Work supported by the Defense Nuclear Agency.

1. Introduction

Pulse power systems employing liquid dielectrics are characterized by time varying electric fields in the range of hundreds of

kilovolts per centimeter. For the larger systems, this corresponds to

megavolt potential differences, maintained over gap lengths of the

order of ten centimeters. Voltage risetimes vary from -£10 ns to

£1 us. A probe is thus required which can measure these large voltages in a frequency range from <300 kHz to £30 MHz.

A simple capacitive E probe can generally be built to satisfy

these requirements. Figure 1 shows a diagram of the device in operation. The probe consists of a small circular portion of the pulseline

outer conductor which has been isolated from the remainder by a 50 Q

impedance. The capacitance, C , of the probe with respect to the

59
pulse! ine inner conductor, together with the 50 ft impedance constitutes an RC differentiation circuit. The resulting signal is thus

proportional to dV/dt and must be integrated, (in our case using a passive RC integrator) to produce a signal proportional to V .

OUTPUT

1 _ . J_ MACHINE (j_

Figure 1. Diagram of E probe.

The bandwidth of the resulting voltmeter is determined by the

requirement that the passive differentiator and integrator truly

differentiate and integrate. This places the following requirement

on the angular frequency co , viz:

In reality, a more stringent limit exists on the high frequency

response. This is caused by a small stray capacitance, (shown in

Figure 1 as C s ), which exists at the cable-pul sel ine interface

and shunts the 50 ft cable impedance to ground. The quantity C s is

estimated from the probe design via electrostatic field plots. By

careful probe construction, the value of C may be pushed well below

25 pF. However, for most practical probe designs C s > Cp and

1/50 ft C

s represents the upper limit on co .

Figure 2 shows the calculated absolute value of probe output

for unit eiwt pulseline voltage. The values for various elements

are chosen to resemble those encountered in the BLACKJACK 5 probes at

Maxwell Laboratories, viz:

C

p

= 0.5 pF

R

i = 5000 ft

C. = 1 nF

60
Various values of Cs are shown, in order to illustrate its effect

upon the high frequency cutoff. The calculation also assumes that the

dielectric is water, with a time constant pe = 120 ys (typical

conditions of BLACKJACK 5). Thus, the water capacitances, Cp

and C s , are shunted by resistances of pe/Cp and pe/Cs respectively. The latter quantity is shunted by 50 Q and is always

negligible. The former quantity affects the probe response at very

low frequencies.

Figure 2. Response of E probe.

The curve in Figure 2 may be divided into four physically distinct regions. Region I, below vL kHz, represents the quasi-DC

limit, where the circuit is acting as a purely resistive voltage

divider with response:

v TM~hQ 50 Q C 7 P robe

= B. = 2.08 x 10" 7 (2)

pulseline pe

In Region II, roughly 10 3

Hz to 10

5 Hz the finite conductivity of

the water is becoming less important, but we are still not satisfying

the first of inequalities (1), i.e., the integrator droop remains

substantial. Region III represents the dynamic range of the probe.

Inequalities (1) are satisfied and the probe response is:

V 50 £7 C a probe = 5 x 1Q -6 (3)

pulseline 1 l

61
Region IV comprises the high frequency limit, where either the

effects of C s are being felt or the second of inequalities (1)

is violated. If we define a five percent decline in probe response as our limit on bandwidth, we see that the upper limit is ^30 MHz for

C s = 25 pF, vLOO MHz for C s = 10 pF and several GHz for the ideal

C s = 0.

2. Probe Calibration - Direct Method

The straightforward method of calibrating this or any voltage

probe is to compare the probe's output with that of some "standard"

voltmeter which measures the same voltage pulse. For a probe with the

sensitivity of ^5 x 10-6 described above, we require a calibration

pulse of vL0 kV in order to provide a substantial output signal

(^50 mV). In addition, the pulse must have dominant frequency components which lie in Region III, the dynamic range of the E probe.

On BLACKJACK 5, this is accomplished with a 20 nF, 100 nH pulser which

injects a pulse on axis at the front of the machine (Figure 3). This

results in a wave of frequency ^3 MHz which travels down the transmission line.

This traveling wave calibration pulse illustrates a general

phenomenon encountered in large systems such as BLACKJACK 5, viz, that

frequencies in Region III correspond to wavelengths shorter than the

size of the system. This means that transit time effects dominate and

it is necessary therefore to place the "comparison" voltage probes

very close to the E probe in order to insure that both are viewing

the same voltage. If we desire instead to have the convenience of

treating the pulse! ine inner conductor as a lumped capacitance with

uniform voltage V(t) , we must limit the bandwidth of V(t) to

regions I and II, and, hence, the output of the E probe will not be

directly comparable to that of a comparison voltmeter.

The BLACKJACK 5 machine has a large number of E probes

along its length and for most of them it is not convenient to connect

up a comparison voltage probe nearby. An exception is provided by the

diode voltage probes at the output end. As shown in Figure 3, these

probes are located just beyond the plastic insulator which separates

the water from the vacuum diode. Thus, by connecting a comparison

probe on the vacuum side close to the insulator it is possible to

sample nearly the same V(t) which is seen by the E probes. We

have used this method of calibration for the diode voltage probes,

using a 1000:1 frequency compensated resistive voltage divider as our comparison probe. The results provide a useful benchmark for evaluating the less direct "capacitance-current" method described in

Section 3.

62
Figure 3. Diagram of direct calibration method.

The attenuation ratio of the resistive voltage probe was checked

using the ±5 volt level sine wave output of the Tektronix SG503 function generator, recently calibrated and traceable to NBS. The probe

output was measured using a Tektronix 485 oscilloscope, calibrated on the 5 mV/div scale and also traceable to NBS. The probe was found to

give nominal 1000:1 division up to frequencies beyond 100 MHz. It was connected across the diode insulator, taking care to minimize the area

of the region bounded by the insulator and the probe leads. This was

to avoid measuring any of the inductive voltage dropped across the

diode and not seen by the E probe.

Figure 4 shows a comparison between the waveforms obtained from

the E probe (with 5.06 ys integrator) and the resistive probe. The

small amplitude ringing on the resistive probe is probably due to

excitation of waves in the resonant structure formed by the probe

leads (^2 m long), which are terminated by the 50 KQ, probe impedance

on one end and the 0.6 Q pulse! ine impedance on the other. The 0.6 Q, end is almost a node for such a system, thus, no ringing is observed

on the F_ probe. The amplitudes of the ringing on the resistive

probe does not constitute a severe noise problem, thus, we can obtain

from Figure 4 the following calibrations for the three E probes:

63
Table 1. Probe Calibrations: Direct Method

PROBE (sens)

-1

= 5.06 vis 50 Q C

P

Derived Cp Value

1 1.903 x 10

5 0.532 pF

2 1.873 x 10

5 0.540 PF

3 1.786 x 10

5 0.567 pF

E PROBE NO. 1 RESISTIVE DIVIDER

0.020 V/div 100 ns/div 2,0 V/div 100 ns/div

E PROBE NO, 2 RESISTIVE DIVIDER

0.020 V/div 100 ns/div 2,0 V/div 100 ns/div

113 I A . • 1

I

E PROBE NO, 3 RESISTIVE DIVIDER

0.020 V/div 100 ns/div 2,0 V/div 100 ns/div

Figure 4, Comparison of E probes and resistive divider.

64
3. Capacitance - Current Method

The E probe is a sufficiently simple and well understood device

that we can state with confidence that its sensitivity (within its

dynamic range) is given by Eq. 3. Thus, calibration of the probe

rests upon measurement of two quantities: the integrator time con- stant, R i C i , and the probe capacitance Cp . Determination of the

former quantity is a straightforward technical task. Some care must

be taken in construction of the integrator in order to insure that it

behaves like an ideal RC integrator out to ^100 MHz. Once this is

accomplished, R-jCi may be measured, for example, by checking the

signal attenuation as a function of frequency and fitting to a [1 + (coRC) 2 ]"'1 expression.

The crucial task is thus measurement of Cp . Since this quantity

can be estimated geometrically to be ^0.5 pf it is too small to be mea- sured with an impedance bridge. To measure it, a method which we call

the "capacitance current" method is adopted. This method uses a slow,

0.99 yF capacitor to inject a current onto the central conductor of

the pulseline. The injected current flows to ground (the outer conductor) largely as displacement current through the pulseline capacitance.

The rest is shunted to ground due to the finite resistivity of water.

A fraction of the displacement current as well as the shunting current

is intercepted by the vol tage .probe and is responsible for the probe

signal. A model of the actual circuit is shown in Figure 5. Cjl

and Rjl are the capacitance and shunting resistance of the pulseline,

(lumped circuit elements on this time scale). Cp and Rp are the

capacitance and resistance between the probe and the central conductor

of the pulseline. Noting that C R = CTI RT, = pe , one can show

that P p TL TL

Figure 5. Model for the voltage calibration circuit.

65
This relationship between C

p

, Cj L, V(t) and I(t) is

independent of the resistivity of water.

For BLACKJACK 5, C-n is approximately 280 nF and, hence, can

be measured accurately with an impedance bridge. I(t) is monitored

with a model 1809 Pearson current transformer and V(t) can in

principle be measured directly with an oscilloscope. However, in

practice a low pass filter is used to filter out high frequency noise

on both V(t) and. I(t) . The 5.06 ys integrator used in actual

operation of the E probe turns out to be convenient for this purpose. As long as the same linear filter is used for both signals, the

ratio of the two waveforms remains unchanged.

Figure 6 shows waveforms obtained for the three diode voltage

probes and the Pearson current transformer. The latter is a selfintegrating Rowgowski coil with a 50 Q output impedance. The R/L

droop rate for the device is quoted at 0.5% ms, negligible for our purposes. The sensitivity of the probe is 0.1 V/A ± 0.1 percent when

unterminated. The same accuracy, and one-half of the sensitivity, is

obtained provided one is careful to terminate in the characteristic

impedance of 50 Q,.

E PROBE NO, 1 E PROBE NO, 2 0,010 V/div 5 //s/div 0,010 V/div 5 ms/div

£ PROBE NO, 3 PEARSON CURRENT PROBE 0,010 V/div 5 ms/div 4.0 V/div 5 ms/div

Figure 6. Waveforms obtained in capacitance-current method.

66
Since the same signal filter must be used on the E probe output

and the Pearson output, it is necessary to rely on shot-to-shot reproducibility, taking repetitive shots at the same charging voltage.

For a given device, it was found that identical charging voltages gave

indistinguishable waveforms, to the accuracy with which oscilloscope

traces can be measured. The charge on the 0.99 yF capacitor was

23 kV for these shots.

From the data in Figure 6, together with a bridge measurement of

Cjl (280.5 nF), one can measure the following values of C and

derive sensitivities based upon a 5.06 ys integrator:

"

Table 2. Probe Calibrations: CC Method

PROBE

c

p VALUE

Derived , c nc „

i

• x.• i \ — 1 5 . 06 ys (sensitivity) " 50 q c

P

1 0.546 pF 1.853 x 10

5

2 0.556 pF 1.853 x 10

5

3 0.571 pF 1.772 x 10

5

4. Comparison of the Two Methods

The values of Cp obtained from the direct and the "capacitancecurrent" methods are in good agreement as illustrated in the following

table:

Table 3. Comparison of Calibrations

PROBE c

p ("C - C")/C ("DIRECT" )

1 1 026

2 1 030

3 1 007

67
Thus, in this case, we find agreement to within three percent

between the direct comparison method and the capacitance-current

method in spite of the highly indirect nature of the latter. This

gives us confidence in applying the capacitance-current method as quick and easy means of calibrating the pulseline voltage probes,

particularly for the less accessible ones.

The authors wish to thank John Shipman for helpful comments.

Work supported by the Defense Nuclear Agency.

6G
MEASUREMENT OF TRANSIENT POTENTIALS

COAXIAL TRANSMISSION LINES USING COAXIAL

INDIVIDERS

N. Fujimoto, S.A. Boggs and R.C. Madge

Research Division

Ontario Hydro

Toronto, Ontario, Canada

Techniques employing coaxial dividers, with flat response from the

kHz region to >1 GHz, have been developed for measurement of transient

phenomena in coaxial transmission lines. These have been used to investigate partial discharge induced phenomena in gas-insulated transmission lines as part of a program to develop incipient fault detection and

location techniques for gas-insulated switchgear. This paper describes

the basic application, the theory and practice of coaxial couplers, and

examples of partial discharge-induced transients.

Key Words: high voltage dividers, partial discharge; transient phenomena.

1. Introduction

A gas-insulated substation consists of a network of coaxial high

voltage conductors, disconnect switches, ground switches, and circuit

breakers all insulated with SF5. For 500-kV switchgear (300 kV normal

conductor-to-sheath potential), the conductor is typically 17 cm and the

sheath 55 cm in diameter. Typical gas pressures range from 350 to

450 kPa. Circuit breakers use SF5 as the breaking medium and are nor- mally based on the "puffer" principle, whereby contact motion forces gas

through the contact assembly to cool and extend the arc. Stations, including circuit breakers, rated to 100 kA fault current are presently

available.

A network of 17 cm aluminum conductors coaxially housed in aluminum enclosures with a lossless dielectric makes a superb high frequency

distribution network, with an attenuation of less than 10 db/km at

1 GHz. These high frequency propagation characteristics facilitate the

characterization and location of high frequency transients within the

69
switchgear. Transients arise from two basic sources, dielectric breakdown and partial discharge.

Dielectric breakdown can occur either line-to-ground, as in the case of system failure (fault), or within the line, as in the operation

of a disconnect switch. In either case, the transient magnitude is substantial and can cause severe overvoltages as a result of doubling at

terminals or the superposition of reflections. System failures related

to switching-induced transients have generated substantial interest in

the subject. As well, these transients, which can have magnitudes in

the MV range with about 7 ns risetimes, cause ground rises in the range

of 100 kV which can disrupt electronic control systems and present a danger to operating personnel [1,2]. Interest in measuring both

internal transients and groundrise transients on external surfaces has

stimulated research into high frequency high voltage measurement

techniques.

Partial discharge can occur from several sources within a gasinsulated substation. Floating components (corona shields, spacer inserts, etc. which are not bonded to the conductor or sheath) can cause

substantial discharge [3]. Free conducting particles can be levitated

in the field and bounce, causing small discharges each time they land

[4]. Filled epoxy spacers are used to support the conductor. A manu- facturing defect, such as a void or stress enhancement on an electrode

can cause discharge within the spacer ("treeing"). Each of these sources of discharge is symptomatic of a defect within the system which can lead to breakdown. Thus, the ability to detect and locate partial discharge sources during commissioning tests and normal service is important to reliable operation. The transient magnitudes generated by these sources of partial discharge have a dynamic range of about 80 db. A

small free-conducting particle generates a signal of about 0.1 V propagating in the coaxial transmission line, while a floating component can easily generate 1000 V transients. Discharge magnitude from "treeing"

varies during the growth process. The transients from all sources of

partial discharge are extremely "fast", with risetimes in the range of

350 ps (the limit of measurement) or less [3].

Under EPRI sponsorship, Ontario Hydro is developing methods for

fault location and partial discharge location which depend on high re- solution timing (fault location) or cross correlation (partial discharge

location) of induced transients [5,6]. These techniques, as well as re- search into the dangers of switching-induced transients, depend on coupling the transients cleanly from the switchgear.

For measurement of external transients (groundrise), a transmission

line divider technique developed by Newi [7] has been used with considerable success. For measurements of transients propagating within the

coaxial switchgear, coaxial couplers have been used. These facilitate

very clean measurement of transients, with measured risetimes limited

only by the 1-GHz 'scope bandwidth over division ratios ranging from 3:1

to 2000:1. The low frequency response can extend to about 10 kHz for a low division ratio coupler to less than 1 Hz for a high division ratio

70
coupler. This excellent performance, extreme versitility, and unparalleled simplicity makes the coaxial divider very valuable for coaxial

systems.

SF6 SYSTEM.

SF6 SYSTEM SHEATH, RADIUS a

SF6 SYSTEM CONDUCTOR,

RADIUS b

ZO

'n-: Ci

j__CCOAXIAL COUPLER

3

CONNECTION TO

MEASUREMENT

DEVICE

COAXIAL COUPLER, RADIUS c Ci = COUPLER - SHEATH CAPACITANCE

C2 = CONDUCTOR - COUPLER CAPACITANCE

Z 0

= SF6 SYSTEM CHARACTERISTIC IMPEDANCE

I] = COUPLER - SHEATH CHARACTERISTIC IMPEDANCE

Z 2 - CONDUCTOR - COUPLER CHARACTERISTIC IMPEDANCE

Z 3 = IMPEDANCE OF EXTERNAL MEASUREMENT DEVICE AND CONNECTION

FIGURE 1 BASIC COAXIAL COUPLER CONFIGURATION

2. The Theory and Practice of Coaxial Capacitive Couplers

Couplers investigated for use in SF5 insulated coaxial systems are thin concentric metal ic cylinders connected to an external potential

measuring device. An infinitely thin concentric cylinder lies on an equipotential surface and does not affect the propagation of signals in

a coaxial transmission

scribed by (Fi gure 1)

:

line. The potential of the cylinder, V

0

, is de

c

1 +c2 z 1+z

2

i c . t a ln

B

+ lric

1 c

(1)

71
The connection between the cylinder and the external measuring device

substantially complicates the analysis of coupler response. The inevitably finite impedance, Z3, causes a high pass characteristic and excitation of coupler resonances. Coupler eccentricity and thickness also cause deviations from the ideal case by introducing impedance mismatches

which lead to signal scattering at both ends of the coupler.

In practice, less than 10% impedance mismatch between the "open"

transmission line and the transmission line in the region of the coupler

is easily achieved through the use of relatively thin couplers, a high

impedance connection from coupler to the measuring device, and reasonable attention to coupler concentricity. The last of these parameters

is least critical and can be evaluated from the equation for the capacitance between two eccentric cylinders [8],

2 7T £

C =

1 n

r , 2 0 (rj

r. , 2

i (r-j

(2)

where r-j and r0 are the radii of inner and outer cylinders, respectively, and x is the distance between their axes. This equation can be used

to calculate characteristic impedances for eccentric couplers, if one assumes that the inductance per unit length does not change appreciably

for small eccentricities. For example, a system with conductor, coupler, and sheath radii of 1.9 cm, 3.8 cm, and 5.1 cm, respectively and a "x" of 0.5 cm will have about 4% mismatch. For this same system (without eccentricity), a 0.3 cm coupler thickness will cause an 8%

mismatch. Figure 2 shows computed step response as a function of the

per cent impedance mismatch. Severe mismatches cause substantial

degradation of coupler risetime.

If the measuring device appears purely resistive, a 50 U terminated

cable, for example, the coupler will exhibit a high pass characteristic

with a response:

_£ = L__ (3)

V. l+jwR(C.+C

2

)

The response at infinite frequency is that of a "perfect" coupler, and

the low frequency response is determined by Ci+C2« By extending coupler

length, low frequency response can be improved. For large coupler

lengths, transmission line effects are superimposed on the lumped parameter high pass characteristic. Figure 3 shows that computed response

72
NO MISMATCH

10% MISMATCH

70% MISMATCH

FIGURE 2 COMPUTER ANALYSIS OF

IRREGULAR COUPLER RESPONSE

HORIZ SCALE: 2 COUPLER LENGTHS/DIV

VERT SCALE: 0.1/DIV, NORMALIZED TO INPUT

I ' ' '

1 1

' ' '

1 i1 1 1 1

i " " i ' i

i iSrrrrjw^- 1

(a)

STEP & PULSE RESPONSE OF 80cm

COUPLER WITH 50ft PICK UP AT

COUPLER'S LEADING END

INPUT: 15V

(b)

COMPUTER ANALYSIS OF

COUPLER WITH 50ft PICK UP

TOP: PICK UP NEAR LEADING

END

LOWER: PICK UP AT COUPLER'S

CENTRE

HORIZ SCALE: 2 LENGTHS/DIV

VERT SCALE: 0.1/DIV,

NORMALIZED TO INPUT

FIGURE 3

COUPLER RESPONSE WITH 50ft PICK UP

73
based on transmission line parameters agrees well with measured re- sponse. The width of the "staircase" is determined by coupler length

and position of the "pickup" along the coupler.

Low frequency response can be extended by many orders of magnitude

through the use of a high impedance "pickup" from the coupler, e.g., a FET probe. The capacitances of such probes (1.5 to 3.5 pF) can cause resonances in coupler response which degrade measurement accuracy. The

precise pattern of resonance-induced errors depends on coupler length

and pickup position. Figure 4 shows the influence of residual coupler

resonance and the effect of pickup position. Frequency doubling when

1ns mi

(a)

PROBE AT LEADING END

OF COUPLER

(b)

PROBE AT COUPLER CENTRE

FIGURE 4 RESIDUAL RESONANCE IN COUPLER RESPONSE

(PROBE CAPACITANCE %2pF)

the pickup is at the coupler center significantly reduces resonance

amplitude, probably as a result of superposition-induced cancellation.

Coupler resonant effects can be reduced by keeping the coupler transit

time less than the risetime of the pulses to be measured. Figure 5 gives an indication of the quality of measurement possible under such

conditions. In practice, coupler length must be based on a compromise

between requirements for low and high frequency response. The coupler

used to record Figure 5a has a -3db high pass characteristic at about

1 kHz. The corresponding figure for Figure 5b is about 10 kHz. Such

cutoffs are not likely to impose serious restrictions for most high

frequency measurements.

74
| 5V

2i

50ns| z

r]

:::: Bzz n zz z

(a)

80 cm LONG COUPLER

INPUT : 100V, 1.5ns RISE

(b)

7.6 cm LONG COUPLER

INPUT : 1 . 5V, 400 ps RISE

THE LARGE NEGATIVE SPIKE AT

10ns IS A REFLECTION FROM THE

END OF THE TESTPIECE AND NOT

ASSOCIATED WITH THE COUPLER

FIGURE 5 REDUCTION OF RESIDUAL RESONANCE

3. Coupler Calibration

Couplers with known geometries really need no calibration. The

calculated response based on known geometry should be within a few per

cent of the measured response. Couplers formed by the application of

metallic foils over insulating tapes on the transmission line sheath re- quire calibration, at least in the form of a low frequency foil -sheath

capacitance measurement. In both cases, response characteristics can be

determined by coupling fast risetime pulses into the transmission line

through tapered conical terminations, which maintain the transmission

line impedance while reducing the diameter to the point, where an appropriate cable connector can be used. If the transmission line impedance

is greater than the generator and cable impedance, a resistor can be incorporated into the tapered coupler. This will cause a reflection back

into the pulse generator, but will provide a matched termination to

reduce ringing within the transmission line during calibration. Such a coupler, with a 10 fi resistor, was used to couple signals from a 50 Q, pulse generator into the 60 Q transmission line used for experiments. For calibration of very high division ratio couplers, high voltage pulse generators are necessary. Generators which will produce 1000

V pulses with 500 ps ristimes at 250 PRF or 200 V pulses with 1 ns risetimes at 10 kHz PRF are commercially available.

75
4. Measurements of Discharge-Induced Transients in Gas-Insulated

Transmission Lines

During research related to the development of discharge location

techniques for gas-insulated switchgear s measurements of transients

typical of a number of discharge sources have been measured using a coaxial dividers shunted by 50 Q, cable and a 1 GHz bandwidth 'scope. A

free conducting particle bouncing in the field generates a very small

transient, in the 100 mV range. The rise time (Figure 6) is probably

limited by the 350 ps ristime of the 'scope and signal degradation in

the connecting cables. A floating component generates a similar transient, except with a magnitude in the 1000 V range (Figure 7).

«Ml

FIGURE 6

TRANSIENT GENERATED BY A FREE CONDUCTING PARTICLE

BOUNCING IN THE COAXIAL TRANSMISSION LINE. THE

MEASURED RISE TIME OF ^500ps IS PROBABLY LIMITED BY

THE LENGTH OF CABLE EMPLOYED. THE DOUBLE PEAK AND

REPETITION AT ^6ns ARE THE RESULT OF REFELCTIONS

WITHIN THE SYSTEM.

FIGURE 7 TRANSIENT GENERATED BY A FLOATING COMPONENT. THE

AMPLITUDE CAN OFTEN BE PREDICTED ON THE BASIS OF

EXPERIMENTAL GEOMETRY.

76
Discharge in a solid dielectric ("treeing") generates very sharp

transients (<350 ps risetime) whose duration and magnitude vary with the

evolution of the tree (Figure 8). In these "treeing" measurements, the

sample was placed at the end of a 60 9, gas-insulated transmission line

with one electrode in contact with the conductor and the other with the

sheath. Any discharges within the sample are "loaded" into the

transmission line, thereby facilitating measurement with the coaxial

divider. The 350 ps risetime measured is probably an order of magnitude

less than can be measured by more conventional means and is limited only

by the 'scope. The true risetime of such phenomena remains to be

determi ned.

MMMHHMM RSImmm msmMM

HRHMKSam IIIHIMBimiti HUH 1MUM HHH MM

FIGURE 8 TRANSIENT GENERATED BY "TREEING" IN A

SOLID DIELECTRIC SPACER LOCATED AT ONE

END OF A GAS-INSULATED TRANSMISSION

LINE. THE RISE TIME MEASURED IS LIMITED

BY THE 1GHz BANDWIDTH OF THE 'SCOPE.

Finally, a high division ratio coaxial divider was used in a study

of the arc current necessary to sustain arcing in a transmission line

with a gap in the conductor. The coupler was fabricated with very thin

mylar tape and thin copper foil over the sheath. A division ratio of

2000:1 was achieved. In Figure 9, one can see the frequency doubling

which occurred when the arc across the gap in the conductor extinguished. The oscillation is "ringing" of the entire testpiece prior to

extinction and half the testpiece after extinction. Figure 10 shows the

breakdown wavefront, which has the 7 ns risetime typical of 60 ft SF5-

insulated transmission line.

5. Conclusion

The measurements of breakdown and partial discharge induced

transients described in this article and the relative ease with which

they are obtained, indicate the usefulness of coaxial couplers for the

measurement of high frequency phenomena in coaxial transmission lines.

77
0

mmmmmwwm

DELAY

FIGURE 9

OSCILLATION OF A TRANSIENT THROUGH AN ARC.

THE ARC ACROSS A GAP IN THE CONDUCTOR ACTS

AS A SHORT CIRCUIT UNTIL NEAR MID PHOTOGRAPH

WHERE A DELAY TO RE-IGNITION IS EVIDENT. THE ARC

EXTINGUISHES ABOUT 1200ns INTO THE PHOTOGRAPH,

WHERE THE OSCILLATION FREQUENCY DOUBLES.

a 2 3kvHlgmm sinmmIB

mmm

il

FIGURE 10

CONDUCTOR GAP BREAKDOWN-INDUCED TRANSIENT.

NOTE THE RISE TIME OF ABOUT 5ns. THE 50ns PULSE

WIDTH IS RELATED TO THE LENGTH OF THE TESTPIECE.

Through the careful design of couplers and ancillary apparatus, the

measurement system can be matched to the specific needs of the experiment for division ratio, frequency response, and freedom from resonances. The use of commercially available FET probes as pickups greatly en- hances low frequency response and facilitates a high pickup impedance

with little attenuation. For many measurements, especially those of

sharp transients or with large division ratios, 50 ft cable can be used

directly from the coupler. Coaxial dividers have and will facilitate

accurate measurements of phenomena which can be "staged" in a transmission line geometry.

78
6. References

Ford, G.L., Boggs, S.A. , and Chu, F.Y., "Transient Groundrise in

SF6 Substations Investigated", Transmission and Distribution , August 1979.

Ford, G.L., Boggs, S.A. , and Dick, E.P., "Transient High Voltage on the Grounded Enclosures of Gas-Insulated Equipment", Proceedings of

the Canadian Electrical Association Meeting , March 1980.

Boggs, S.A., Ford, G.L., and Madge, R.C., "Coupling Devices for the

Detection of Partial Discharges in Gas-Insulated Switchgear",

Paper, IEEE PES Winter Meeting, February 1980.

Wooton, R.E., "Investigation of High Voltage Particle-Initiated

Breakdown in Gas-Insulated Systems", EPRI Research Report EL-1007,

March 1979.

Boggs, S.A. , Ford, G.L., and Chu, F.Y., "Partial Discharge Location

in Gas-Insulated Switchgear", Gaseous Discharges II,

L. Christophorou, Ed. Pergamon Press, New York, 1980.

Boggs, S.A., Chu, F.Y., and Ford, G.L., "Improved Fault Location

for Gas-Insulated Switchgear", Transmission and Distribution , July

1980.

Newi , G. , "A High-Impedance, Nanosecond Rise-Time Probe for Measuring High-Voltage Impulses", IEEE Transaction , PAS87 , 9, 1779

(1968).

Finkelman, M. , "The Variation of the Capacitance of Cylindrical

Electrodes Due to Loss of Concentricity", Ontario Hydro Research

Division Report No. E66-43-K, September 1966.

79
A VOLTAGE MONITOR FOR MAGNETICALLY INSULATED TRANSMISSION

LINES*

Regan W. Stinnett

Sandia National Laboratories

Albuquerque, NM 87185

Magnetically insulated transmission lines pose special

problems in voltage measurement because of the extremely

high electrical stresses that are typical in them and

because of the initial electron loss which is intrinsic to

the establishment of magnetic insulation. This new voltage

monitoring technique uses negative ions which have been

recently discovered in magnetically insulated transmission

lines. I u -* These negative ions are produced in the cathode plasma and are accelerated by the gap voltage toward

the anode. If two positively biased collectors are placed

at different distances behind a hole in the anode, these

negative ions may be collected to give time resolved signals

which will provide the data necessary for a time of flight

unfold of the gap voltage. A small magnetic field applied

to the flight path can be used to deflect any electrons

produced in the gap while not significantly deflecting the

much heavier negative ion.

Although a complete voltage unfold may be difficult in

practice, this technique may be used to obtain peak gap

voltage very simply. If the collectors are located far

enough away from the source, then the first negative ions

to arrive will be those with the highest velocity, hence

those produced at peak gap voltage. The velocity, and

energy of these negative ions may be obtained to a few percent accuracy from arrival times at the two collectors.

1. Introduction

Magnetically insulated transmission lines (MITL's) are

of interest in mult iter rawatt accelerators used in simulation and particle beam fusion. '2 ,3] These transmission

"*"Triis wor"E~was supported by the U. S. Department of Energy,

under Contract DE-AC04-76-DP00789

.

80
lines achieve power densities as high as 10 J watts per

square centimeter because they operate at electric stress

levels of over 10 MV/cm.

4 J This is possible because in a MITL electrons are prevented from crossing the vacuum gap

between the cathode and the anode by a transverse magnetic

field which causes them to move in cycloidal orbits. When

the magnetic field is strong enough the electrons turn around

before reaching the anode and magnetic insulation is

achieved. This is shown schematically in Figure 1.

MAGNETICALLY INSULATED FLOW

Figure 1. Magnetically insulated flow

Because of the high electric stress and the electron

orbits in the vacuum gap between the anode and cathode it

is difficult to directly measure the voltage in magnetically

insulated transmission lines although such a measurement is

extremely important. A method for measuring this voltage

has been discovered which employs negative ions produced

in the MITL's.

2. Negative Ion Formation

In MITL's a cathode plasma is formed by explosive

emission ' 5 ' 6 1 when the electric field on the cathode rises

81
above 250 kV/cm. The constituents of this cathode plasma

are not well known but seem to include both electrode

material and hydrocarbon contaminants on the cathode

surface A'* The electron density is in the range of 10

cm

-3 ^-"with a plasma temperature of a few eV.'- ' Explosive emission and desorption of gas molecules from the cathode could provide up to 10 6 neutral atoms and molecules per

square centimeter. These conditions, together with the

excess of electrons in the cathode plasma resulting from

the nature of magnetic insulation, provide an ideal situation

for the creation of negative ions. An examination of the

relevant cross sections and processes involved in negative

low production in MITL's will be the subject of a later

report

Because the negative ions are produced in the cathode

plasma they are accelerated to the full gap voltage as they

move across the vacuum feed gap, so their energy accurately

reflects the gap voltage at the time they crossed the gap.

There are several atoms and molecules, such as H, C, 0, and

which have bound negative states. Different negative

ion species produced in the cathode plasma would arrive at

a collector at different times, producing distinct peaks,

which could be used to cross check the gap voltage.

3. The Experiment

Van Devender, Stinnett, and Anderson recently discovered the presence of negative ions in the long magnetically insulated transmission lines of Mite and Hydramite,

modular prototypes of Sandia National Laboratories Particle

Beam Fusion Accelerator PBFA I. Both of these accelerators

produce a 2 MV, 400 kA pulse lasting 35 ns. This pulse

travels down a 6 meter long, magnetically insulated triplate

transmission line which is 20 centimeters wide and has a

1 centimeter gap. Both the anode and cathode surfaces are

coated with a 25 micron thick graphite coating. These

MITL's have been demonstrated to operate efficiently '11-'

on both the MITE and Hydramite accelerators.

Further experiments on Hydramite have been performed

to study negative ion production in greater detail. In

these experiments a time of flight spectrometer (TOFS),

shown schematically in Figure 2, was mounted behind a 0.635

cm hole in the anode and used to collect the negative ion

current. The first collector was a 50% transmitting screen,

allowing the second collector, mounted 40 cm behind the

first, to collect negative ions along the same flight path.

Both collectors were biased positively at 300 volts to

recapture any secondary electrons emitted. This arrange82
80 cm

Figure 2. Time-of-f 1 ight spectrometer used in Hydramite

negative ion experiments

ment made it possible to measure the negative ion velocities

directly. The collectors were mounted at distances from

15 to 81 centimeters behind the gap.

Several techniques were used to confirm that the negative signals from the collectors were actually caused by

negative ions. A magnetic deflector was mounted on the TOFS

to sweep electrons out of the flight path. This magnetic

deflector produced a transverse magnetic field of 500 Gauss

over a distance of 10 cm. When a 70 micron thick aluminum

foil (much less than one electron range but more than one

ion range) was placed in the flight path betwen the MITL

and the first collector, the collector signals were reduced

by an order of magnitude, verifying that the signal was

mainly due to negative ions rather than electrons.

4. Experimental Results and Analysis

The TOFS was used to measure voltage on the Hydramite

MITL on several shots. Figure 3 shows the signals from

collectors located 41 and 81 cm from the cathode on Hydramite shot 8936. The typical voltage at the insulator is

shown in Figure 4. The dotted line shows the expected

voltage waveform 5 meters down the MITL where the TOFS

was located. The erosion and steepening of the leading

edge of the pulse is caused by the lossy region at the

front of a magnetic insulation wave and is typical.

Negative ions leaving the cathode plasma cross the

vacuum feed gap in less than 3 ns so it is valid to assume

a constant voltage to calculate their time of flight across

the gap. The time of flight to the collectors is determined

33
40 cm:

80 CM: - -

Figure 4. Typical voltage at the input of the Hydramite

MITL. The dotted line indicates the expected

waveform 5 meters into the MITL.

84
by the same gap voltage. Negative ions formed on the rising

portion of the voltage waveform will experience "bunching"

as the higher energy ions catch up to those formed at

lower voltage. If the flight path is sufficiently long

the negative ions formed at peak voltage will arrive at the

collectors first, making it simple to determine peak voltage.

This occurs on Hydramite with the collectors located at 41

and 81 cm for negative ions heavier than hydrogen. This

technique also works .for shorter flight paths if the approximate voltage waveshape is known so that small corrections

can be made to the arrival times based on an estimate of the

bunching. It should also be possible to determine V(t) by

locating collectors near the anode where bunching is a small effect. This technique has not yet been used on

the Hydramite negative ion data.

The data in Figure 3 from shot 8936 can be used to

illustrate the technique for determining peak gap voltage.

The leading edges of the first peaks on the two collectors

are separated by 24 ns. These first peaks are presumably

due to H~ . Using the voltage waveform shown in Figure 4,

one finds that the H~ ions created at peak voltage should

arrive first of the collector located at 81 cm, but would

arrive about 4 ns later than the leading edge at the 41 cm

collector. It is now possible to estimate the velocity

of the H~ ions to be 2 cm/ns with an accuracy of about

+ 10%. This implies a peak gap voltage of 2 MV + .4 MV.

"Dsing this voltage one can then identify with reasonable

confidence the second peak as being due to C~ . This is

reasonable since both electrodes are coated with carbon.

These second peaks are very sharp because of bunching.

Again using the voltage waveform of Figure 4, one finds

that the C~ ions created at peak voltage should arrive first

at both collectors. The leading edges of the second peaks

are separated by 68 + 2 ns. THis gives a peak voltage

of 2.1 MV + .1 MV. The next significant peak in the

collector signals can now be tentatively identified as C

2

based on this peak voltage.

5. Conclusions

These experiments provide the first detailed measurements of negative ions produced in magnetically insulated

transmission lines. Several species including H~ , C~, and

C

2 ~ have been tentatively identified. These negative ions

are accelerated to the full gap voltage in MITL's and

provide an excellent, non-perturbing diagnostic of peak

voltage which should be accurate to better than 5%.

35
These negative ions may also have important implications in MITL loss processes. In addition the negative ions

produced in MITL's may have possible applications as a source

of high energy neutral beams and in a negative ion diode.

These experiments have demonstrated the use of negative

ions as a voltage monitor. The techniques employed are both

simple and general. Confidence levels and simplicity can

be further increased by a definite determination of the

negative ion species. This determination was done by

Stanley and Stinnett and is being presented in another

paper at this conference.

6. References

[1] T. H. Martin, Proceedings International Pulsed Power

Conference, 1976, paper ID-1.

[2] E. I. Baranchikev, A. V. Gordeev, V. D. Korolev, and

V. P. Smirnov, Sov. Phys. - Tech. Phys. 3, 42 (1977).

[3] S. Shope, J. W. Poukey, K. D. Bergeron, D. H. McDaniel,

A. J. Toepfer, and J. P. VanDevender, J. Appl . Phys.

49 (7) , July 1978.

[4] G. Dahlbacka, C. Gilman, C. Meins, H. Helava, I. Roth,

R. Schneider, Bulletin of the Am. Phys. Soc. 25 (8),

October 1980, paper 1E3

.

[5] S. P. Bugaev, E. A. Litvinov, G. A. Mesyats, and

D. I. Preskurovskii , Sov. Phys. -Usp., Vol. 18, No. 1,

p. 51.

[6] G. N. Fursei and V. M. Zhukov, Sov. Phys. Tech. Phys.,

Vol. 21, No. 2, February 1976, p. 176.

[7] Baksht, Bugaer, Keshelev, Mesyats, Stas'ev, Sukhushin,

and Timofeev, Sov. Tech. Phys. Lett. 3 (7), July 1977.

[8] D. W. Swain, S. A. Goldstein, L. P. Mix, J. G. Kelly,

and G. R. Hadley, J. Appl. Phys. 48 (3), March 1977.

[9] R. B. Baksht, B. A. Kablambaev, G. T. Razdobarin,

and N. A. Ratakhin, Sov. Phys. Tech. Phys. 2_4 (6),

June 1979, p. 689.

[10] Negative Ion Losses in Magnetically Insulated Vacuum

Gap, J. P. VanDevender, R. W. Stinnett, and R. J.

Anderson, J. Appl. Phys. to be published February 1980.

[11] J. P. VanDevender, J. Appl. Phys. 50 (6), June 1979.

86
MEASUREMENT OF MAGNETICALLY INSULATED LINE VOLTAGE USING

A THOMSON PARABOLA CHARGED PARTICLE ANALYSER

Timothy D. Stan ley

Diagnostics Division 4243

Regan W. Stinnett

High Density Plasma Source Division 4234

Sand i a National Laboratories

Al buquerque, NM 87185

Abstract

The absence of direct measurements of magnetically insulated line

voltage necessitated rel iapvce on inferred voltages based on theoretical

calculation and current measurements. This paper presents some of the

first direct measurements of magnetically insulated transmission line

peak voltages. These measurements were made on the Sandia National

Laboratories HydraMITE facility III.

The peak voltage is measured- by observing the energy of negative

ions produced at the line cathode and accelerated through the line

voltage. The ion energy and the -charge- to-mass ratio are measured

using the Thomson Parabola mass spectrometry technique. This technique

uses parallel E and B fields to deflect the ions. The deflected ions are detected using a microchannel plate coupled to a phosphor screen

and photographic film.

The Thomson Parabola results are compared to Faraday Cup measurements and to calculated voltages based on current measurements. In addition, the significance of observed positive ions will be discussed.

Key Words: Magnetic Insulating Voltage Measurement, Negative Ions

Thomson Parabola Charged Particle Analyser

1 . I ntroduct i on

Magnetically I nsu I ated Transm i ss i on Lines (MITL) [2] are an essential component of light ion beam fusion devices because they permit

many radial pulse power lines to converge to give very large power

densities. Voltage measurement of MITLs is critical to establishing

37
energy transport and power available at diodes for particle beam fusion.

Measuring the voltage on MITLs is difficult because conventional

monitors, i.e., capacitors or resistive voltage dividers, disrupt electron flow causing the magnetic insulation to break down. In the absence

of d i rect vo I tage measurements, a rather complex theoretical realationship between measurable currents and the line voltage was established by

C. W. Mendel [31. Because the sensitivity of the inferred voltage to

small time shifts in the currents, a more direct voltage measurement was needed. This paper presents the theoretical basis and results for a MITL voltage measurement technique based on measurements made with a Thomson Parabola charged particle analyzer.

2. Theoretical Basis

R. W„ Stinnett and J. P. Vandevender [4] observed that during operation of a MITL, negative ions are produced at the cathode and are accelerated through the line potential to the anode. Measuring the peak

energy of the ions exiting through a hole in the anode gives the peak

line voltage. The Thomson Parabola charged particle analyser was used

to determine the ion energy as wel I as the ion species.

2.1 Thomson Parabola Charged Particle Analyser (TPCPA)

The TPCPA operates by deflecting a col I imated ion beam in

parallel E and B fields [5]. The E field produces a vertical deflection

proportional to the ion charge state divided by the ion kinetic energy

while the B field produces a horizontal deflection proportional to the

charge state divided by the ion momentum. Therefore, ions with an energy

distribution trace out a parabola of the form yxx^ with the peak energy

determined by the point of least deflection on the parabolas (i.e., the

point on the curve closest to the neutral particle spot formed by un- detected neutral particles and photons). Each unique ion specie and

charge state produces a separate parabola. The deflected ion current is

amplified by a microchannel plate (CEMA) and then activates a phosphor

screen to expose a photographic film. Figure 1 gives the definitions

and equations used in describing the TPCPA in this paper.

2.2 TPCPA Accuracy

The TPCPA used in these measurements is described in detail in

Reference 5. The deflections in mm for this instrument are given by

equations (1) and (2). The terms in these equations are defined in F i gure 1

Y = C

£

V Z/E (1)

X = C

B BZ/y EA (2)

88
Y (mm) - Ce_VZ

CE VA X

2

(CBB) 2 z'

CB BZ

REFERENCE

COORDINATE SYSTEM

(ORIGIN CENTERED ON NEUTRAL

PARTICLE SPOT)

ACTUAL

COORDINATE SYSTEM

NEUTRAL

PARTICLE SPOT

E

X (mm) = E = ION KINETIC ENERGY (keV) v fc A A = ION ATOMIC MASS (AMU)

Z = ION CHARGE STATE

B = MAGNETIC FIELD (489 GAUSS)

V = VOLTAGE ACROSS DEFLECTION PLATES (4000 VOLTS)

CE

= ELECTROSTATIC DEFLECTION CONSTANT (1.561)* Cb= MAGNETIC DEFLECTION CONSTANT (.8475)*

*THESE CONSTANTS ARE DETERMINED BY INTERNAL

DEVICE GEOMETRY

Figure 1. Thomson Parabola equations and definitions

For this instrument, the charge to mass ratio can be determined with an uncertainty of + \\% white the velocity can be determined to within

+ 7%. Solving equation (1) for E/Z yields:

This quantity, E/Z, can be determined to within +_ ~b% [51.

Voltage measurements were made on the Sandia National Laboratories'

(SNLA) HydraMITE facility. This facility has a nominal operating

voltage of 1.9 MV at 0.5 MA current in a 40 nsec pulse. The ions for

voltage measurement were extracted through a port located midway along

the MITL (see Figure 2). The B sensors for calculating the line voltage were located at the diode end of the MITL [61. A Faraday cup to obtain

line voltage from the time of flight of the ions was also located on the

diode end of the MITL. Most of the data for this paper was obtained

during a series of nine shots conducted on December 10-12, 1980. For

these shots, the MITL was terminated with a razor blade diode, and the

HydraMITE operation was nominally the same.

E/Z = C

E V/Y (3)

3. Experimental Setup

89
Figure 2. Experimental setup

4. Results

Figure 3 shows the negative ion parabolas for the first shot of

this series. It is typical of the parabolas from this series. Figure 4

is a computer simulation of. parabolas for H~, C~ , 0~ , and ions

with energies from 200 keV to 2 MeV using the equations (1) and (2).

From comparison of the photo with the simulation, it is apparent that

the left-most curve is H . Some ambiguity exists for the remaining

curves, for example, 1 MeV 0" ions and 1 MeV CF^ ions would have the

same Z/E and Z/P. Also, the closeness of the curves makes resolution

somewhat difficult. However, the other three distinct curves are most

likely C~, 0" and • Note that the perpendicular lines on the

photograph are for reference between photographs and not the coordinate

system for measuring curves. The origin for the coordinate system is

the center of the neutral particle or photon-generated spot in the lower

righthand corner of the photograph which is about 2 mm left and 1 mm above the intersection of the reference lines.

The voltages measured in these shots are summarized in Table 1. The Faraday cup voltage comes from a t ime-of-f I i ght unfold of Faraday

cup collected ion current to determine negative ion energy [61. The

current-derived voltage is obtained from boundary currents measured with

small pick-up loops (B probes) and then applying Mendel's model 131.

Several current derived voltages are missiag due to a loss of current

data. Much of the fluctuation in these voltages results from difficulties in correcting for time shifts in the cathode and anode current

measurements. The electrostatic voltage was measured from the vertical

deflection of the maximum energy points on the Thomson Parabola photographs while the magnetic deflection measurement was determined by the

horizontal de-flection to the maximum energy poi-nt of the hydrogen curves

Shot 9068 was used to obtain a horizontal reference thus providing no voltage information while Shot 9074-was used for a vertical reference

and therefore only provided an electrostatic deflection measurement.

90
Figure 3. Negative ion parabolas from shot 9067*

Z/P (mm)

Figure 4. Computer simulation of Thomson Parabola curves*

*The scale sizes on the photo and computer simulation were changed in

reproduction. The horizontal reference line in the photo was originally

57 mm long while the horizontal axis in the computer simulation was 60 mm

long.

91
Table 1. Peak voltage measurement summary

SHOT

NUMBER

ELECTROSTATIC*

VOLTAGE (MV)

MAGNET I Ct

VOLTAGE (MV)

CURRENT DERIVED

VOLTAGE

FARADAY CUP

VOLTAGE (MV)

9067 1 .8 1 .6 2.2 2.1

9068

9069 1 .9 1 .9

9070 1 .9 1 .6 1 .6

9071 1 .6 1 .5 2.1 1 .5

9072 1.8 1 .6

9073 1 .6 1 .4 1 .7

9074 1 .7 1 .8 2.0

9075 2.1 1 .7 1 .4 1 .4

Ave 1 .8 + .2 1 .6 + .2 1 .8 + .3 1 .7 + 3

*From equation ( 1 )

.

tFrom equation (2) using H~ curve.

The small deflection in the photographs from these shots (approximately

4 mm in the vertical direction and 11 mm in the horizontal direction)

compared to the width of the lines and the neutral particle reference

spot severely limits the voltage resolution on these tests. Since these

tests, two improvements have been made to the system that increase the

resolution by at least a factor of four. First aluminum wedges were

placed over the deflection plates. These wedges increase the electrostatic field thereby more than doubling the ion deflection while not

perturbing the B field. By cross calibrating between electrostatic and

magnetic deflections, the electrostatic deflection constant (Cc) in

equation (1) was found to increase from 1.56 to approximately 4.0.

Scheduled calibration on a Van de Graaff accelerator w-i I I identify this

constant to within 3 percent.

The second improvement uses the observation that positive ions at

peak line voltage also exist. We hypothesize the positive ions are produced from the negative ions through a strippin-g process during

transit to the analyser which preserves the peak energy. This hypothesis

is supported by the observat i on—that shortening the drift region before

deflection reduces the intensity of the positive ion traces. Through

centering the parabola display, both positive and negative ions can be

observed simultaneously. Then the deflection can be measured between

positive and negative ions of the same species, doubling the measured

deflection and eliminating the need to use the large neutral particle

spot as a reference.

Using the Pythagorean relationship and the parabolic relationship

between X and Y, a single equation giving the ion energy as a function

of the radial distance from the neutral particle spot can be obtained.

92
This relationship is:

2 CE

V Z

- K + K

2 + 4r2

(4)

where r is the radial distance from the neutral particle spot to the

maximum energy point on the parabola and

(CB

) 2 B

2 Z

K ' CE VA * (5 »

Using this relationship, the peak voltage can be measured with

approximately 30 mm deflection between the peak energy point of the H+ and H~ ions with line widths less than one mm, thus permitting

relative voltage variations as sma I I as + 5% to be resolved. Figure 5 shows the centered display with both positive and negative ions

(negative ions are deflected upward) and the increased vertical deflection resulting from the aluminum wedges over the deflection plates.

Figure 5. Parabolas from HydraMITE shot 9117

93
5 . Cone I us i ons

An accurate technique has been developed to measure magnetically

insulated line peak voltages with a Thomson parabola particle analyzer.

Experimental results show that measured voltages are in reasonable

agreement with calculated peak voltages deduced from boundary current

measurements and time of flight measurements.

Acknow I edgement

The authors thank Glenn W. Kuswa for his interest and suggestions,

John Anthes for providing the Thomson Parabola, Carl Cianciabella for

his technical support, James Chang for helpful suggestions, and the

Sandia HydraMITE Operations staff.

6. References

[I] "Particle Beam Fusion Research Facilities at Sandia National

Laboratories," Pulsed Energy Directorate, Sandia National

Laboratories, 7/80.

[21 J. P. VanDevender, "Long Se I f-Magnet i ca I I y Insulated Power Transport Experiments," J. Appl. Physics 50(6), June 1979.

[31 C. W. Mendel, Jr., J. P. VanDevender, and G. W. Kuswa, "Determination of Line Voltage in Se I f-Magnet i ca I I y Insulated Flows." IEEE

Second International Pulsed Power Conf., Lubbock, TX, June 12-14,

1979.

[4] J. P. VanDevender, R. W. Stinnett and R. J. Anderson, "Negative Ion

Losses in Magnetically Insulated Vacuum Gaps," accepted for publication in Applied Physics Letters.

15] M. A. Gusinow, M. M. Dillon, G. J. Lockwood, L. E. Ruggles, "The

Thomson Parabola Ion Analyser, SAND78-0336, March 1978.

[61 R. W. Stinnett, "A Voltage Monitor for Magnetically Insulated Trans

mission Lines," Workshop on Measurement of Electrical Quantities in

Pulsed Power Systems, Boulder, CO, March 2-4, 1981.

8. Sponsor Acknowledgment

This work supported by the U. S. Department of Energy under

contract DEAC04-76-DP00789.

94
Discussion Session 2

Terry McComb (NRC, Canada). If I may just consider our previous

discussion where we were talking about the appropriateness of the

two high voltage divider systems. Basically, these dividers are being used to calibrate the efficiency of lasers and the interest is

to make voltage measurements to be combined with current measurements

to look at the input power to a laser and compare this with the actual

lasing power. The sort of accuracy that we have for the measurement

of the power input from the laser is in the order of 10%. So, basically,

we are looking at 5% for both the current and voltage accuracy. But,

as I pointed out, the present accuracies are not at that level yet

and we are still working toward those.

Marco Pi Capua (Physics International). I would like to make a couple

of remarks on voltage measurements with liquid dielectric resistor

dividers. The first one addresses one of the comments of

Richard Malewski. Richard's remarks about the dependence of the

dielectric constant on frequency are correct. The way that

Physics International has built dividers over the years is to

incorporate the tap-off within a homogeneous column. In this way,

the non-linear effects that exist in the dielectric resistive fluid

and the variations of resistance and dielectric constants with

temperature and frequency are compensated because the divider is a geometric divider. A second remark is that this type of divider will

work in vacuum when the divider vessel is assembled from field grading

rings and cone insulators. We have described such a monitor which

operates up to two megavolts in a vacuum. We have used this to measure voltage directly in magnetically insulated transmission lines

by connecting the divider to the inner conductor of the transmission line

through a hole in the outer conductor. We measured voltages up to one megavolt with penetrations in the transmission line which are

1 1/2 inches in diameter. A sketch of the voltage divider appears in

figure 1 below. This type of measurement was described in the

literature [1].

Jack Katzenstein (Maxwell Laboratories). I have a comment on the

paper by Dr. Stanley. Let me say I think that it is a very elegant

application of what is also a very elegant technique, the Thompson

parabola mass spectrometer. But, there is a more elegant, a more

recent mass spectrographs technique which would be pertinent, namely

the Bainbridge velocity selector. The deflection in the Bainbridge

Mwo-Megavolt Divider for Pulsed High Voltages in Vacuum, D. Pellinen

and M. S. Di Capua, Rev. Sci . Inst. 51, 1980, pp. 70-73.

95
TO(-)HIGH

V VOLTAGE

Plastic

Metal

Resistive Solution

Figure 1. Sketch of vacuum voltage divider. Metal annuli and

plastic cone frustra form containment vessel for resistive

solution. Inductance of the measuring circuit Li places a lower bound on divider resistance. Stray electron current

places upper bound on resistance. The strays C$ and C$

9 h are much smaller than the interstage capacitances. Divider

is geometric so RjC^ = R2C2*

96
velocity selector is proportional only to the velocity of the ion and

independent of its species. It was used by Bainbridge as a device

prior to magnetic deflection to sort the ions according to the charge mass ratios. It occurred to me that it would be easier to interpret

the deflection produced by the Bainbridge velocity selector which uses crossed E and B fields instead of parallel fields then by using the

parabol ic method.

Tim Stanley (Sandia). I guess I have to plead ignorance on the

Bainbridge velocity selector so I will have to look that up.

Jim Chang (Sandia). I think that an appropriate response to this

question is that we do have multiple species and would like to make a survey of the species component of this negative ion beam at the same time as determining the energy (the peak energy in this case). We

are working on a time resolution capability by sweeping the electric

fields so that we do get essentially the time resolved voltage

measurements of the magnetically-insulated transmission lines.

Mark Wilkinson (Maxwell Laboratories). While we are on the topic of

this negative ion measurement, I would like to ask what current

densities were observed for the negative ion beams.

Regan Stinnett (Sandia). It depends on the preparation of the lines.

It seems to be typically one or two amps per square centimeter. We

coat our lines with about 25 microns of carbon to form a lot of

whiskers so that you have a uniform cathode plasma. We find

that on the first shot, after coating the line with carbon, you get a much larger loss than that, but it then settles down to something on the order of a couple of amps per square centimeter. That is total

current including all of the different species.

Pi Capua . I want to know what the lag time is between the onset of

voltage on the line onset of negative ion emission.

Sti nnett . As far as I can tell it is immediate and there seem to be more negative ions produced at higher voltage or higher current. We

are running matched so they occur at the same time.

Frank Young (NRL). I have a question for Tim Stanley. If you look

at the parabolas produced by the Thompson parabola probe, you see some structure running from the high energy point further up the

parabola to lower energy down along the parabola. This is particularly

evident for the hydrogen line. What information can you extract from

this structure?

Stanl ey . If you remember from the talk, there was a little bit of

disagreement between the data and the computer simulation. If you

look at the original of this photograph very carefully, you will see that there is a line which continues up past that discontinuity.

Now, it turns out my apertures are a quarter-inch hole in front and

97
then a small 20-mil pin hole in back. Because of my device geometry,

I can have an eighth of an inch difference in my line positions. It

turns out that my lines don't fill up that total area and, therefore,

it appears that the cathode plasma is not continuous but that there

is emission from discrete points. I think I am seeing discontinuities

across the cathode.

Neville Harris (Ion Physics). I noted that you had positive ions and

this is presumably due to the poor vacuum. If you can get reactions

to actually remove two electrons from the ion, can you also not get a change of charge during flight, so that the velocity does not correspond

to the complete voltage falling across the line? Do you get any

aperture effects such as bombardment at the edge of the aperture with

secondary stuff coming off the edges?

Stanl ey . I have a separate vacuum system where the deflection region

is. The poor vacuum is in the region of the tube that connects my

Thompson parabola apparatus to the magnetic insulated line. It is

pumped from within the magnetic line through a quarter-inch hole. I don't think I am seeing any stripping or charge exchange after I get

through my aperture and into my deflection regions. The current is

fairly low at the second aperture. Something like 10^ - lOl 2 ions

actually are exposing my film. I don't think that I am seeing any

slit effects here. Other effects that may have been produced on the

front aperture or space charge expansion are restricted by the second

aperture.

Richard Malewski (IREQ). In the second part of the presentation,

there was a description of the Ontario Hydro device for measuring

voltage in a gas-insulated bus which is exactly the case as we referred to

in the first part of this discussion. The difference is the dielectric.

According to Neville Harris, there is some damping provided in the

circuit which starts at high voltage bus and which ends at the ground

envelop, due to the resistance of water. If we refer to a simple

RLC circuit with this capacitance, which seems to be predominant

with some inductance at, let's put it this way, a resistance, if I remember correctly, the critical damping resistance should be less

than 1/2 /L/C which gives us about 200 ohms. I don't believe that

the water used here will have resistance less than 200 ohms. So, I don't believe that water will act effectively as a damping element.

I would rather stick to the first statement that this is an integrating

element which simply does not pass the high frequency oscillation.

The second comment is concerning the Ontario Hydro system, where

there is a tapping device between the coaxial bus and ground. Depending

on the length of the tapping sector, one can get more or less smooth

response of such a divider. A suggestion is to not make it out of

aluminum but make it out of a resistive material such as a resistive

paint on some insulating substance. That will provide you with a regularly shaped response because you will damp internal oscillations

in this sector. Try this, it should work. I agree with Mr. Di Capua

98
that that is a very clever trick to tap the liquid divider

column here, and it will certainly help to give the response that

isn't oddly shaped. But that's the first approximation effect. If

you really look at the equivalent circuit of such a divider, there

will be a high voltage conductor, the resistive elements of this

liquid, and the parallel capacitors. These elements do vary in the

same way in the upper and lower parts of the divider. But, don't

forget that we have also the ground here and we have stray capacitance

to ground as well as stray capacitance to the high voltage bus.

Ideally, the voltage distribution along the divider column should be

such that the capacitive voltage distribution is the same as by

resistive distribution. Unfortunately, this is very difficult to

obtain because we cannot control the stray capacitance. It depends

on the relative position of the divider relative to the bus high

voltage. Since we have in the same circuit frequency and temperature

variable elements, as well as stray capacitance which is not temperature

nor frequency dependent, we might expect second order problems.

Pi Capua . I agree that one has to consider the stray capacitance

between the divider and ground and the divider and the high voltage

conductor. We have built a device in which the interstage capacitance

in the liquid dielectric is much larger than the strays to ground or the strays to the high voltage conductor. The high capacitance arises

from the water dielectric. The strays are shown schematically in the

voltage divider sketch (figure 1). Another remark is that the

measurement gets difficult when transit times within the divider

itself which are of the order of the duration of the pulse you are trying to measure. In this case, we try to position the divider so that the wave which is the voltage we are attempting to measure is normally

incident to the divider column. Even then the going is sort of

rough. The question then becomes really what are you measuring and I don't intend to enter into that discussion now.

Chat Cooke (MIT). One difference between the two capacitive dividers

that Harris and Fujimoto presented is that the electric fields are oriented differently in the two. In one case, Harris', the low

voltage capacitor is actually perpendicular to the main gap capacitance

and field, while in the second case, Fujimoto's, the low voltage

capacitor is in fact geometrically parallel to the main gap capacitance

and hence the fields are in the same direction. My question is:

besides convenience, is there any advantage or disadvantage between

those two arrangements?

Harris . I will first deal with the point that Richard drew with

regard to the damping of this circuit. I am saying that the series

resistor which goes to ground at the oscilloscope termination forms a damping element for the oscillatory circuit in the divider. Now,

back to the question raised by Chat Cooke. I have the low voltage

capacitor perpendicular to the high voltage. I was working in a very

high voltage line where the operation was near breakdown. If you

have the capacitors parallel, then fundamentally the average field in

99
the two capacitors must be the same. If, due to mechanical errors,

the sensing element is at a slight angle, the field at the end is

going to be greater. The field will also be increased by the insertion

(of low dielectric constant) of the monitor cable. In addition,

there is the field enhancement in the corners. However, if you recess

the sensing electrode (see figure la of Discussion Session 1) you no longer get this trouble. By simply making the area of this cylinder

(corresponding to Cg) greater than the area of the end of the plug

(corresponding to C\ ) , you can lower the field in the recessed region.

The other advantage is that of mechanics. When you wish to put it in

a line, you can insert the complete assembly from the outside. If you

have a mushroom shape, you are going to have to get to it from both

sides. So there is mechanical convenience as well. That's the reason

for the perpendicularity of the low voltage capacitor. Also note that

because you have lowered the field in the low voltage region, you have

lowered the requirements for mechanical tolerance with regard to your

device. With regard to the monitor described by Mr. Fujimoto and the

problems due to transit time between the ends of the sensing electrode,

there is a very simple way to deal with that. You taper it. You don't

have it the same length all the way around the circumference. Then

transit time varies from location to location around the tube.

John Shipman (Sachs-Freeman, NRL). In a voltage divider circuit such

as Neville Harris described, one can better understand the nature of

the coupling to the divider by considering a short impulse wave traveling down a coaxial transmission line with the divider mounted

on the outer conductor. The impulse wave will be accompanied by a thin disc of displacement current (and conduction current, if the

medium has conductivity) flowing radially between the inner and outer

conductor. Since there is no inductance associated with this thin

disc, the coupling to the divider will be capacitance and resistance

in parallel only and will occur as the disc passes over the divider.

In regard to ringing or resonance of such divider circuits, I did have a high frequency resonance at about 2500 MHz with my dividers

using Mylar between the center cylindrical plug and the outer grounded

cylinder. This resonance occurs because the plug and outer cylinder

form a low impedance, short transmission line essentially open

circuited at both ends. The resonant frequency is that at which the

line is a half wavelength. In Mr. Harris' case with water between

the plug and the grounded wall, if the plug were 3 inches long, the

resonant frequency would be 220 MHz and would only be visible on very

high frequency oscilloscopes if excited by very fast rising pulses.

N. Fujimoto (Ontario Hydro). Dr. C. Cooke raised a question regarding

the orientation of the low voltage capacitor with respect to the high

voltage capacitance in two designs of capacitive dividers. Initially,

capacitances involved were considered to be lumped. However, presently

our preference is to use a transmission line analysis where the

coupling cylinder defines two transmission line impedances Zi and Z2

100
for a short length within a coaxial system of impedance Zg (figure 1 of our paper). This type of analysis illustrates the prime advantage

of this geometric configuration. As described in the paper, if the

sens'ing cylinder is perfectly thin and concentric, then no impedance

mismatches are formed at either end of the coupler. As a result,

any propagating wave (assumed to be TEM) incident onto the coupling

system does not suffer any degradation in its high frequency characteristics and the division ratio is simply determined by the ratio of the

transmission line impedances. Indeed we have been able to measure transients with risetimes about 350 ps (the limit of our ability to

make single shot measurements) as shown in figure 8 of our paper.

Degradations that do occur are felt to be caused by the connection

to the external measurement device, which because of stray capacitance,

may appear to have initially a low impedance. Admittedly, some disadvantages in this configuration may exist in high voltage

applications. In practice, we would normally install a shunt inductor

on the low voltage portion of the divider for 60 Hz protection and

flare the coupler ends to prevent corona.

The suggestions made by Messrs. Malewski and Harris concerning

the use of resistive couplers and the use of tapered ends on the

coupler are interesting, but I will have to withhold comment until I have the time to examine these suggestions in more detail.

John Rarth (Barth Electronics). When you drive the output into a 50 ohm 1 i ne, you could have a directional coupler which would be the

secondary transmission line B where the outputs would come out through

the walls. As long as the secondary line (B) was terminated in its

characteristic impedance, you would not have any resonances on it.

You would have a short time response, of course, but you do get an exact division ratio, and you would have as fast a response as the

loss factor of the insulating material would permit.

Bob Hebner (NBS). I think that in the last few minutes of the

discussion, I am going to invoke my prerogative as Chairman and try

to elicit a bit of discussion on a few of the other topics which were presented earlier, but which we have not had any discussion on so far. For example, we were treated, at the opening of this session,

to a very eloquent discussion of electro-optic measurements of voltage

by Jim Thompson. I was wondering if anyone had any questions or comments about that presentation or that field.

Harri s . I noticed that you said that the number of fringes was proportional to the square of the voltage. You really mean the square

of the field, and this is not just a nitpicking point because there

are end effects in a Kerr cell. In other words, as the light goes

through the cell, it goes through end regions as well, which have

much lower fields and thus don't respond in the same way as the main

field. Is that not true that you are going to have to compensate for

that in a rather mathematical way?

101
Jim Thompson (Univ. of South Carolina). Yes, that is true. But,

usually what you assume though is the Kerr cell length is long compared

to the path length through interaction of the fringing fields. That

is the way one usually says that this is negligible. Now, I think

that Bob Hebner has done some work and seen that this is a few percent

or one percent or less for reasonably long Kerr cells. If one wanted

to make a very short Kerr cell to reduce the capacitance, for example,

then that would be a problem.

Hebner . When we are developing high accuracy Kerr cells, we calibrate

them by measuring what we call the cell constant. This is done

by applying a known voltage and measuring the phase shift induced

by that known voltage. This calibrates both the uniform field and

the end field effects which range from maybe a half percent to

two or three percent. We find that the end field effects are only

geometry dependent and, therefore, having performed a voltage

calibration, you have an end field effect which is a constant

forever and you can use the cell for extended periods with very

high accuracy. Over the past twenty years, techniques have been

developed to account for this adequately so that measurements to

about 0.1% precision can be made. We think that by the end of the

year we will see about 1/4% accuracy.

Katzenstein . I don't think one ought to dismiss the Pockels effect

out of hand for electro-optic voltage measurements for two important

reasons. It is a longitudinal effect, and it is a linear effect.

Since it is a longitudinal effect, it should be possible to eliminate

a geometric factor in the calibration such that only the Pockels

constant enters. The fact that it is linear is going to greatly

facilitate a numerical inversion. Of course, I must agree with

Dr. Thompson that Pockels active materials are expensive and they

have somewhat questionable dielectric strengths.

Malewski . I have had experience with an application of the Pockels

effect to high voltage measurement. The problem was to measure

voltage distribution along the column of a lightning arrestor under

impulse conditions. We devised a simple gadget composed of a heliumneon laser whose beam was deflected by two roof prisms, passed through

a Pockels cell, and was returned to the laser beam detector standing

on the ground some distance from the high voltage object. We were

able to measure the voltage applied to the Pockels cell which was installed in sequence at each particular shed of the lightning

arrestor. The system worked fairly well but the main problem was ringing. Every electro-optic crystal is at the same time piezoelectric

which means that when you apply a voltage to it, it is stressed and

the mechanical response for such a rapid stress is an oscillation.

We tried several tricks. Prestressing techniques used to eliminate

this mechanical ringing were not too successful. A typical oscillogram

showed a signal due to the high voltage impulse we applied perturbed

102
by one due to the mechanical vibration of the slab of crystal which

was excited by the electric field applied to the metallized transparent

pi ates

Katzenstei

: It becomes a matter of time scale. The acoustical

vibration of a crystal is going to be on a several microsecond time

scale. If you are measuring voltages on nanoseconds or several

nanoseconds time scale, that's important.

Gene Nolting (NSWC). I would like to ask a question of both

Bob Hebner and Jim Thompson. I found the method of using the laser

to make high voltage measurements extremely interesting. However, I have some trouble relating how to use this in a real high voltage

system and it appears perhaps that maybe the best way to incorporate

this technique would be to calibrate monitors which are a little more mechanically easy to use on existing machines. Is it possible for

you to, say, tap off the high voltage somehow and calibrate existing

monitors with this technique? Perhaps you could have a traveling

road show going around the country and making measurements so that

everybody's measuring voltage with the same standard.

Hebner . I will comment first. Your point describes a process

that we have used in the measurement of microsecond and longer

pulses. One can use the Kerr effect devices to accurately

transfer among laboratories. I think that a classic example

is that for about ten years, when Sandia was having some interesting activities with a certain waveshape in the microsecond

regime, accurate voltage measurements of those pulses were maintained

among NBS, Sandia, and a variety of their subcontractors using

transportable Kerr effect standards. That is an approach which has

shown itself to be useful over a limited regime of voltages and time.

In addition, your suggestion to use a technique that has totally

different sources of error from a conventional probe as a comparison

to determine information about the accuracies of the existing probes

is a good one. In our program at NBS, we have adopted exactly that

phi 1 osophy

Thompson . As far as calibration is concerned, I think that Bob's

comments are correct. I think in the laboratory if you have to measure voltage, you reach for a capacitive or resistive voltage

divider thinking that it is going to be easy. Then, you spend several

months trying to get rid of your loops, noise, and various other

things. It may be that we should begin to use electro-optical

techniques more than we do if for nothing else than to see waveshapes.

103
Session B - Power and Energy Measurements

ION CURRENT AND VOLTAGE DETERMINATIONS

BY NUCLEAR TECHNIQUES

Frank C. Young

Plasma Physics Division

Naval Research Laboratory

Washington, DC 2037 5

Abstract

Nuclear reactions induced by intense proton or deuteron beams from

high-power pulsed generators can be used for current and voltage determinations. Average currents and voltages are determined from measurements of delayed radioactivity from nuclear reactions. This technique

is illustrated for proton current using the * C(p,y) N^yeactio^gnd for

deuteron current and energy using the * C(d,n)* N and AH(d,p) Ail re- actions. Time-resolved current and voltage determinations can be made

from measurements of prompt nuclear radiations. Proton currery^s are determined by prompt-gamma measurements using the F(p,ay) 0 re- action. Neutron time-of- flight techniques are described to determine

either the deuteron energy or current using the d(d,n) He reaction.

*1. Introduction

Pulsed power generators are being used to produce intense pulsed

ion beams, and new techniques are required to determine the energy and

current of such beams [1] . The ions are usually produced in these generators in a diode region at high vacuum where it is difficult to make

electrical measurements. The current and voltage from the generator can

be measured reliably across an insulator in an oil or water dielectric

just before the vacuum region. The electrical characteristics in the vacuum diode may differ from those measured in the dielectric. An

inductive correction (-L dl/dt) is usually applied to the voltage where

L is the inductance that the pulse "sees" in the vacuum region and I is

the total diode current. Electron emission from surfaces in the vacuum

other than the cathode causes the diode current to differ from the

current measured in the dielectric. Also this electron emission makes

it impossible to determine L for the inductive voltage correction.

Consequently, the electrical properties measured in the dielectric are

not necessarily the same in the diode. In this presentation I shall

review some new diagnostics based on nuclear techniques that have been

104
developed to measure the energy (i.e. voltage) and current of proton or deuteron beams.

Many of the usual techniques for ion-beam measurements are inappropriate for the beams from pulsed power generators. These beams are frequently so intense that scintillators, for example, do not survive

the burst of ions. Furthermore, these beams are generally space-charge

neutralized so that charge collection and magnetic or electrostatic

deflection are complicated and uncertain. In addition, a beam pulse may

contain several different species (e.g. H, D, C,....).

The most unambiguous method for measuring a specific beam component

in this situation is by selective nuclear reactions. Information con- cerning the energy and intensity of specific ion species can be inferred

from measurements of nuclear reaction products. The number of nuclear

reactions N with cross section a produced by M particles of energy E

incident on a target of thickness Ax with n target nuclei/cm is given

by N = MnoAx. The dependence of N on energy per particle is contained

in a provided Ax is sufficiently small that a does not change appreciably as the particle traverses the target. The dependence of N on intensity (i.e. current) is contained in M. In general these dependences on energy and intensity can not be separated readily to provide

independent measurements of ion voltage and current. In a few specific

cases this separation is possible. More often, an ion current and

voltage deduced in some other way are used to calculate the expected

number of nuclear-reaction products for comparison with experiment.

Differences are then ascribed to an incorrect current or voltage. These

may then be modified appropriately to bring the calculated number of

nuclear-reaction products into agreement with experiment.

2. Radioactivity Measurements

The technique of nuclear activation has been used to provide information concerning the energy or intensity of proton or deuteron beams.

The approach is to measure the delayed gamma-ray activity induced in a target by the beam. Target samples activated by the beam are removed

and counted with a shielded gamma-ray detector. By using specific

targets and measuring specific nuclear reactions, the energy or intensity of a specific ion species is determined.

The following steps are taken to optimize the sensitivity of the

activation. First, thick targets are used to optimize the number of

reactions. In this case, the reaction yield is given by the integral of

the cross section over the range of the ions in the target. The yield

may be written

Y = N/M = / a/e dE (1)

where e is the stopping cross section of the target. The stopping cross section is proportional to the stopping power ( e = dE/dx) and has

105
been tabulated in the literature [2] for protons and deuterons. Second,

short-lived activities are used. If a small number of radioactive

nuclei (N) are produced, then a short lifetime (t) is desired to

increase the activity (i.e. dN/dt = N/t). However, the lifetime must

be long enough to allow the target sample to be removed for counting.

Half-lives of the order of minutes are convenient. Third, good detection efficiency with adequate gamma-ray energy resolution is provided by

Nal detectors. To achieve low background the detectors may be shielded

with a few inches of lead. Also, for positron activities coincidence

counting with two detectors at 180° to each other is used to reduce the

background by several orders-of-magnitude.

The intensity of MeV proton beams may be determined by using (p,y)

resonance reactions [1]. The cross section for such a reaction peaks at

the resonance energy ER

. For a resonance with a narrow width T, the

reaction yield is nearly constant provided the proton energy E is well

above the resonance energy (i.e.(E-Ep) >> T). This situation is illustrated in Figure 1 for the 12c(p,y) N reaction [3]. This reaction has

a resonance at 0.46 MeV (T=37 keV) and a

0.2 06 1.0 1.4 1.8 2 2 PROTON ENERGY (MeV)

Figure 1. Thick-target yield for the C(p,y) N reaction in a carbon

target.

resonance at 1.70 MeV (T=67 keV^. j_ Between these resonances the yield

is nearly constant (Y = 7.5x10 N/proton at 1 MeV). In this energy

range the number of protons is given by N = M/Y where M is the number of

radioactive N nuclei produced by the reaction. The 10-min N radioactivity can be measured efficiently with very little background by

coincidence counting of the annihilation^ ^radiation from the positron

decay of N. In this way, as few as 2*10 protons of 0.6 to 1.5 MeV

energy can be detected. For an accurate determination of the intensity

106
of protons greater than 0.8 MeV, it may be necessary to correct for the

C(d,n) N contribution to the N activity from the naturally-occurring isotopic abundance of deuterium in the ion source [ l]

.

Nuclear activations can be used to determine the ion energy or voltage because the yields of most nuclear reactions are strongly dependent on the ion energy. For example, the yields of deuteron-induced

reactions are small below the Coulomb barrier but increase rapidly ft with

energy as illustrated in Fig. 2 for the 12C(d,n) 13N and AH(d,p) AH

reactions [A]. These reactions do not have narrow resonances, but they

CO

~z. o

or

UJ

hZD

UJQ

toO

cr

UJ

Q_Q

_J

UJ

>-o

10'

10

CJS io-'

or

10

/l2 C(d,n) l3 N /

V /^7Ai(d,p) 28Ai

/N— - RATI °

1 1 / i i i

10'

I0

J >

<\l\Q

o

2 y

10

0 12 3 4

DEUTERON ENERGY (MeV)

12 13 Figure 2. Thick-target yields fq^ the ^d,n) N reaction in a carbon

target and for the A£(d,p) AH reaction in an aluminum

target. Ratios of these two yields are also shown.

do reach yields about 10 times larger than (p,y) reactions. These

reactions produce short-lived delayed gamma-ray activities which are

readily measured with a Nal detector [A] . Simultaneous activations of

carbon and aluminum can be used to determine mean deuteron energies.

The ratio of the yields of these two reactions is a s^ns^give function

of deuteron energy (see Figure 2) so that a measured N/ AH ratio can

be used to determine the mean deuteron energy. Then the reaction yield

at this energy is used to determine the number of deuterons.

107
These nuclear activation techniques are not practical for ion beams

of high current density. Ablation of the surface of the target being

activated limits the technique to current densities of about 100 A/cm

depending on the target material, ion species and energy. This limit can be extended an order-of -magnitude by using attenuating screens to

reduce the beam intensity or a "pillbox" to confine the ablated radioactivity [5] . Even so, current and voltage determinations from radioactivity measurements correspond to values averaged over the duration of

the pulse and can only be compared with average values of the electrically measured current or voltage.

3. Time-Resolved Measurements

The prompt-gamma and neutron time-of- flight (TOF) techniques provide time-resolved measurements which allow more detailed comparisons

with the electrical measurements. Also, these measurements of prompt

nuclear radiations are uneffected by ablation of the target.

3.1 Prompt-Gamma Technique

The prompt-y technique is used to measure proton current [6] . A

nuclear react ion^which produces prompt y rays of several MeV energy is

selected. The F(p,ay) 0 reaction, which produces y rays of 6.1 to

7.1 MeV, has been used with Teflon targets. These y rays are measured

with a scintillation-photomultiplier detector of good time resolution

( <5 ns FWHM) . The detector is located in a lead shield to attenuate

the bremsstrahlung ( < 1 MeV) from the generator and to convert

the prompt y radiation into a secondary spectrum of lower energy quanta

which can be detected efficiently with a small plastic scintillator.

The voltage output V^ of the prompt-y detector (prompt-y response)

depends on the proton current I and the nuclear reaction yield Y which

is a function of the proton energy E. To determine the proton Current

from the measured prompt-y response, one must correct for the flight

times of different energy protons from the diode to the Teflon target.

Then the time variation of V-, is related to the time variation of I in d p the following way:

00 V

d

(t

d

) = K / I (t) Y [ E ( t ) ] 6[t

d-t-A/v(t)] dt. (2) o

The distance I is the proton flight path from the diode to the target,

and v is the velocity of protons of energy E = tyg

Dv^ emitted at time t from the diode. The thick-target yield Y (E) for this reaction has been

determined [7] , The quantity K is a constant specified by the detector

geometry and target-to-detector distance, and it can be determined by

calibration with a Van de Graaff accelerator [7]. In practice, the

proton current is determined by iterating on Eq. 2 until the prompt-Y

response calculated with an assumed proton current agrees with the

108
measured prompt-y response. An initial proton current may be determined

from a Rogowski coil measurement [8] of the total ion current or from an

analytic model [9] of ion generation.

An application [7] of the prompt-y diagnostic is illustrated in

Figure 3. The prompt-y response measured from a Teflon target located

60 cm from an ion diode is shown in Figure 3a. The prompt-y signal is

delayed in time due to the proton time-of-flight so that the diode

brems strahlung precedes the prompt-y signal. The measured response is

compared with the response calculated using the proton energy and

current displayed in Figure 3b. The proton energy is given by the

measured diode voltage corrected for inductive effects. The width of

the prompt-y response 'is directly related to the width of the current

pulse. The shape of the prompt-y response reflects the shape of the

proton current in the vicinity of its peak.

o- 6

LJO

o

5

o

o

(E

Q.

a

T

>00

a

PROMPT GAMMA

0

1°\o BREMSSTRAHLUNG To / \ - \J V

A hk t

1 1 "^is

0 20 40 60 80 100 TIME (nsec)

120 140 160 60 80 100 TIME (nsec)

Figure 3. The measured prompt y-ray detector response (solid line) in

(a) is compared with the response (open points) calculated

using the proton energy and current displayed in (b)

.

The proton current determined by the prompt-y measurement is com- pared with the proton intensity determined from the activation of carbon

in the Teflon target. For this measurement, attenuating screens were

used between the diode and the target to minimize target ablation. The

ion energy is small enough that no correction for deuterons in the

proton beam is necessary. A^ m^surement of the positron decay of N

indicates that (5.9±0.6)xl0 N nuclei were produced in the Teflon

target. The number of nuclei produced by ^the proton current and

voltage displayed in Figure 3b is (4 . 7±0 .7 )xlO . These independent

determinations are in agreement within the experimental uncertanties.lt

should be noted that the prompt-y measurement only samples the current

above 340 keV and that the carbon activation only samples the current

above ~ 440 keV. These measurements are insensitive to the current

profile for proton energies below these thresholds.

109
3.2 Neutron Time-of-Flight

Hie pulsed nature of these ion beams makes the time-of-flight (TOF)

technique ideal for measuring neutron energies. The neutron energy is

determined from a measurement of the neutron flight time for a known

flight path. Then the ion energy (i.e. voltage) is determined from the

kinematics of the neutron-producing reaction. With fast scintillators

and photomultipliers a detector resolution of about 5 ns can be

achieved. For neutrons of MeV energies flight paths of several meters

are used to separate the neutron pulse from the diode bremsstrahlung.

As the flight path (A = vt) is increased, the neutron energy resolution

scales inversely with the flight path for a fixed time resolution

6t(i.e 5E/E = 2v6t/£), but the number of neutrons incident on a given

detector decreases quadratically with distance. A compromise must be

made between signal intensity and energy resolution.

The time spread of the neutron response in a TOF detector is

composed of three contributions: l)the detector response, 2)the energy

dispersion of the neutrons and 3) the duration of emission of the

neutrons. For many plasma sources the detector response time can be

made much less than the other contributions. Ion-energy measurements

require a long neutron flight path so that the time spread from energy

dispersion dominates the duration of neutron emission. For ion-current

determinations, a short flight path is required so that the time spread

from energy dispersion is minimized relative to the duration of neutron

emission.

These techniques are illustrated with some measurements [10] using

the d(d,n) He reaction. Kinematics [11] for this reaction are shown in

Figure 4. The neutron energy depends most sensitively on the deuteron

energy for neutrons emitted in the forward direction. Therefore TOF

detectors should be deployed in this direction for ion-energy measurements. The neutron energy is almost insensitive to the deuteron energy

near 90°, and this direction may be more useful for deuteron-current

measurements

.

A determination of the energy and emission time of the most

energetic deuterons from an ion diode using the TOF technique is

presented in Figure 5. The deuteron beam is directed onto a thick

deuterated polyethylene (CD 2) target to produce neutrons by the

d(d,n) He reaction. The responses of two neutron TOF detectors at 15°

and at two different flight paths are shown in Figure 5a. Both

detectors record an initial pulse resulting from diode bremsstrahlung

before the neutron responses are measured. The response corresponding

to the longer flight path is delayed as expected. The energy and

emission time of the most energetic neutrons are deduced from these

traces. The most energetic neutrons have a flight time corresponding to

the leading edge of the neutron response. (This time is measured at

one-half of the signal maximum.) The neutron flight time for each

detector is measured relative to the peak of the x-ray signal and

corrected for the x-ray flight time. The difference of the neutron

flight times implies a neutron energy of 3.41+0.15 MeV. The corre110
120°

2 1 I

I

1 1 0.2 0.5 1.0 1.5 2.0 DEUTERON ENERGY ( MeV)

4. Dependence of the outgoing neutron energy at different angles

on the incident deuteron energy for the d(d,n) 3He reaction.

X-RAYS Q

1 \-—NEUTRONS

1.91 m

1 ^-Jl i i i i i

7.42m

X-RAYS

f

NEUTRONS^/

\

\.. 1 1

i i 1 100 200

TIME (nsec)

300 400

>

2 500

1 400

w 300

2 200

100

0

EMISSION TIME

I

40 60 80

TIME (nsec)

120

ire 5. (a) Neutron TOF measurement for two detectors at 15° to the

deuteron beam direction and at flight paths of 1.91 m and

7.42 m respectively. The maximum deuteron energy corresponds

to the most energetic neutrons which are detected at the

leading edge of the TOF traces. (b) The emission time of

these deuterons is compared with the deuteron energy trace.

Ill
spending deuteron energy is 0.45±0.10 MeV based on the d(d,n) He

reaction at 15°. The uncertainties in these values are a result of

uncertainties in time and distance measurements. The neutron flight

time versus distance is extrapolated to zero distance to determine the

emission time for the most energetic neutrons.

The emission time and deuteron energy are compared with the time

dependent accelerating voltage in Figure 5b. The ion energy trace in

this figure corresponds to the measured diode voltage corrected for the

diode and drift tube inductance and reduced by the energy loss of the

deuterons in traversing a thin foil used for current neutralization

[8] . The maximum deuteron energy for this trace is 0.46 MeV in good

agreement with the energy determination by TOF. The neutron emission

time is detemined relative to this ion energy trace through a fiducial

timing pulse. This corresponds to a time of 67 ns on the ion energy

trace in Figure 5b. This time must be reduced by the flight time of

deuterons from the diode to the target (20 ns) to give the emission time

of deuterons from the diode of 47±7 ns as shown in Figure 5b. The

uncertainty in this value results from propagating uncertainties in time

and distance measurements. The deuteron emission time corresponds to

the maximum deuteron energy within experimental error.

In this example the neutron energy was measured to a precision of

about 4% corresponding to a deuteron-energy precision of about 20%. The

deuteron-energy precision improves at higher deuteron energy because the

deuteron energy is increasing more rapidly than the neutron energy (see

Figure 4). For example, a 4% neutron energy precision corresponds to a deuteron-energy precision of about 10% for 1.5-MeV deuterons.

The neutron response in a TOF detector provides a direct measure of

the deuteron current if the width of the response results primarily from

the duration of neutron emission. The measured neutron response can be

compared with a calculated response to determine the deuteron current.

For an instantaneous pulse of deuterons N^(t) of energy E^ at time tn incident on a thick target, the neutron response is given by

E,

R (t ) = N,(t) f

a (o/e)dE = N,(t) Y (E.) (3)

c n d ^ d n d

where t = t + % /v + H^/v,, n n n d d

I = neutron flight path

v„ = neutron velocity

n

J E, =V?m,v, = incident deuteron energy

3 a = cross section for the d(d,n) He reaction

e = stopping cross section for deuterons in polyethylene.

The neutron response is related to the deuteron current through the

reaction cross section integrated over the target thickness. It is

112
assumed that the target thickness is larger than the range of the

deuterons. The time of arrival of a neutron at the TOF detector (t ) is

different from the time that the deuteron was emitted from the diode by

the flight time of the deuteron to the target (^j/v,) and by the neutron

flight time (£ /v ). The slowing-down time or the deuterons in the

n n target is negligible compared to the neutron flight time. Since the

deuterons lose energy in the target, the outgoing neutrons have a velocity dispersion and hence a time dispersion. In addition, an angular

dispersion of the deuterons incident on the target (e.g. in a focusing

geometry) will produce a time dispersion through the reaction kinematics

(see Figure A). For this discussion we will assume a unidirectional

beam on target. Complications arising from angular dispersion are discussed elsewhere [10]

.

o Cross sections for the d(d,n) J He reaction have been well documented

[11], and the dependence of the neutron yield Y

n on the deuteron energy

is shown in Figure 6 for two different angles. At all angles Y

DEUTERON ENERGY (MeV)

Figure 6. Thick-target yields for the neutron output from deuteron

bombardment of a CD£ target. Yield curves at 15° and 45° are displayed for the d(d,n) He reaction (dashed curves) and for

the sum of the d(d,n) He and 12 C(d,n) N reactions (solid

curves) .

increases with energy. Consequently, the neutron response is largest

for the most energetic neutrons and decreases in magnitude at later

times due to deuteron energy loss in the target, ^,Also shown in Figure 6

is the total yield including neutrons from the C(d,n) N reaction on carbon in CD 2* These neutrons are separated in time from the d-d

neutrons, but they must be included in total yield measurements with

neutron activation detectors [12] . Such total yield measurements can

113
provide an absolute calibration of the deuteron current. Cross sections

for the C(d,n) N reaction are taken from the work of Lefevre [13].

In an experiment the deuteron energy is a time-dependent pulse of

finite duration. Also the neutron yield depends on the angle of the

neutron defector (6 ) relative to the deuteron-beam direction. Then

Eq. 3 must be rewritten to include the effects:

£d (t)

Rnnn^ (t ,9 ) = f N,(t) Y[E.(t)] 6(t -t-* /v -i/v, ) dt (4)

d d nnndd

The neutron velocity is determined by the kinematics of the d(d,n) He

reaction for given values of E^ and 9 . A straightforward inversion of

Eq. 4 for deuteron current N^Ct) is not possible. Instead, the right

side of Eq. 4 is calculated and compared with the measured neutron

response. For this purpose, the corrected diode voltage trace is used

for E^Ct) and the deuteron current predicted by a simple analytic model

[9] is used for N

d (t)

.

Neutron TOF measurements to determine the deuteron current from an

ion diode are presented in Figure 7. Detectors are located at 15° and

85° to the incident-beam direction, and short flight paths are used to

minimize the neutron time spread from energy variations. However, the

flight paths used are sufficiently long for the detectors to recover

from the x-ray flash before recording neutrons. In this figure measured

TOF responses are compared with yields calculated according to Eq. 4.

The deuteron energy E^Ct) and the deuteron current N^Ct) for these

calculations are given in Figure 7a. Neutron yields are calculated in

2-ns time increments of E^Ct) and N^Ct) and then summed to give the curves which are plotted as open points in Figures 7b and 7c. The

calculations have been normalized to the peaks of the measured

responses. The widths of the calculated and measured responses are in

good agreement, but there are some differences in shape. The doublehump in the calculated responses is due to this same structure in the

energy. The energy profile can be modified to bring the calculated

responses into better agreement with experiment. Finally, the energy

and current profiles can be combined with the thick-target yield to

calculate the time-integrated neutron output. The result is normalized

to the neutron yield measured with an activation detector to set the

absolute scale of the deuteron current in Figure 7a.

To assess the importance of the duration of the deuteron current

and energy traces on the neutron response, a calculation was carried out

using a monoenergetic deuteron pulse of constant current of only 2-ns

duration. Neutron responses calculated in this case are shown at the

bottom of Figures 7b and 7c. Here the temporal dispersions at both

angles are much less than the contribution from the duration of ion

emission as given by the ion current and energy traces in Figure 7a.

114
9

TIME (nsec) TIME (nsec)

Figure 7. Comparison of neutron TOF traces with calculations using the

energy and current traces displayed in (a). The measured TOF

traces (solid curves) are compared with calculated yields

(open points) for detectors at (b) 1.91 m and 15° and at (c)

2.36 m and 85°. The results of calculations for a monoenergetic deuteron pulse of 2-ns duration are shown at

the bottom of (b) and (c)

.

115
4. Summary

Currents and voltages determined by nuclear techniques are consistent with electrical measurements. The generator voltage with inductive

corrections and the current from a Rogowski coil can be used to reproduce measured prompt-y and neutron TOF responses. The precision of such

measurements is about 20%.

There are many nuclear reactions which can be used for current and

voltage measurements. In this report only a few have been presented to

illustrate different diagnostics techniques. Other reactions ^hat have

been used include: l)delayed radioactivity from the N(p,y) 0 reaction to measure the current of photons greater than 280 keV [1],

2) prompt-y measurements with the Li(p,y) Be reaction to determine

proton current [14], and 3)neutron TOF with the Li(p,n) Be reaction to

measure the energy of protons greater than 2 MeV [15] . Many potential

reactions remain to be exploited, particularly if reactions producing

delayed (3-radioactivity are considered.

New developments and improvements in these techniques can be

expected as more attention is devoted to using nuclear measurements to

understand the behavior of current and voltage in the vacuum diode

region. The emphasis to date has been on characterizing the ion beam

rather than relating the nuclear measurements to the electrical characteristics of the generator. Nuclear techniques will become even more

attractive as pulsed-power technology moves toward higher voltage

because the increased reaction yields will allow ever smaller ion

currents to be measured.

5. Acknowledgements

The encouragement of D. Mosher, G. Cooperstein, S. A. Goldstein and

J. Golden in the development of these diagnostics is acknowledged.

Also, discussions with J. Shipman were helpful in the preparation of

this report.

116
6. References

[I] F. C. Young, J. Golden and C. A. Kapetanakos, Rev. Sci. Instrum.

_48, 432 (1977).

[2] H. H. Andersen and J. F. Ziegler, The stopping and Ranges of Ions

In Matter , Vol. 3 (Pergamon, New York, 1977).

[3] J. D. Seagrave, Phys. Rev. _84, 1219 (1951).

[4] F. C. Young and M. Friedman, J. Appl. Phys. 46_, 2001 (1975).

[5] A. E. Blaugrund and S. J. Stephanakis, Rev. Sci. Instrum. 49 , 866

(1978).

[6] J. Golden. R. A. Mahaffey, J. A. Pasour, F. C. Young and C. A.

Kapetanakos, Rev. Sci. Instrum. 49_, 1384 (1978).

[7] F. C. Young, W. F. Oliphant, S. J. Stephanakis and A. K. Knudson,

IEEE Trans. Plasma Sci. PS-9, 24 (1981).

[8] G. Cooperstein, Shyke A. Goldstein, D. Mosher, R. J. Barker, J. R. Boiler, D. G. Colombant, A. Drobot, R. A. Meger, W. F. Oliphant, P. F. Ottinger, F. L. Sandel, S. J. Stephanakis and F.

C. Young, "NRL Light Ion Beam Research for Inertial Confinement

Fusion," Fifth Workshop on Laser Interaction and Related Plasma

Phenomena, Rochester, New York, Nov. 5-9, 1979.

[9] S. J. Stephanakis, D. Mosher, G. Cooperstein, J. R. Boiler, J. Golden, and S. A. Goldstein, Phys., Rev. Lett. _3_7.» 1543 (1976).

[10] F. C. Young, D. Mosher, S. J. Stephanakis, S. Goldst ein and D. Hinshelwood, NRL Memorandum Report 3823 (Washington, DC, August

1978).

[II] H. Liskien and A. Paulsen, Nuclear Data Tables JJL_, 569 (1973).

[12] F. C. Young IEEE Trans. Nucl. Sci. NS-22 , 718 (1975).

[13] H. W. Lefevre, C. A. Burke and R. M. Bahnsen, University of

Oregon Report RLO-1925-74, 1971 (unpublished).

[14] R. J. Leeper, E. J. T. Burns, D. J. Johnson and W. M. McMurtry,

Workshop on Measurement of Electrical Quantities in Pulsed Power

Systems, Boulder, CO, March 2-4, 1981.

[15] R. A. Meger, F. C. Young, A. T. Drobot, G. Cooperstein, S. A.

Goldstein, D. Mosher, S. E. Graybill, G. A. Huttlin, K. G. Kerris

and A. G. Stewart, NRL Memorandum Report 4477 (Washington, DC,

March 1981) .

7. Sponsor Acknowledgements

This work was supported in part by the Defense Nuclear Agency and

in part by the Department of Energy.

117
ELECTRICAL MEASUREMENT TECHNIQUES USED AT THE CASINO FACILITY

E. NOLTING, R. MARTIN AND M. RUPPALT

Electronics Hardening Branch

Naval Surface Weapons Center

White Oak, Silver Spring, Maryland 20910

The Casino Facility is a Defense Nuclear Agency nuclear weapons

effects simulator. The machine consists of four-one megavolt fieldemission diodes each capable of producing a one-half terawatt power

pulse. The diodes are located on the end of four separate, 4 MV

water-dielectric coaxial lines and can be energized singularly or in

combination. Each diode has its own set of diagnostic monitors.

Transient diagnostics include voltage monitors, on the pulse-forming

lines adjacent to the vacuum diodes and current sensors, on each of the

diodes. Normal shot-to-shot variation in the Casino output necessitates

accurate measurement of voltage and current with computer aided data

reduction to determine diode performance. This paper describes the

relevant features of the hardware and software used in the Casino

digital data acquisition system. Techniques used to calibrate the

various diagnostic monitors are presented including a discussion of the

procedure used to time phase the current and voltage signals. This is

followed by a comparison of the calculated delivered energy with

total absorbing electron beam calorimeter measurements.

1. Introduction

The Defense Nuclear Agency's Casino Facility was built to produce

bremsstrahl ung radiation for the study of its effects on military

weapons systems and components. In order to produce this radiation

the simulator generates intense, relativistic electron beams that are directed to a x-ray converter. Figure 1 is an electrical schematic of

one-half of the simulator which shows that it is generically similar to

other single-pulse, high-power machines used in inertial confinement

research, material response studies, etc. Each of the two, 58-stage,

4 MV, 400 kJ, marx generators are connnected to two, 13 nF» waterdielectric, coaxial, pulse-forming lines (PFLs). The voltage on each

PFL rises to a maximum of four megavolts in approximately 1.5 ysec,

after which time the energy is transferred by water switches

to matched impedance lines. These coaxial transmission lines are tapered to reduce their output impedance by one-half and this causes

118
TAPERED TRANSMISSION LINE

MARX GENERATOR -

Figure 1. Schematic diagram of one-half of the Casino nuclear

weapons effects simulator. The marx generator applies

four million volts to the pulse forming lines which is

reduced, after switching and transforming to one million

volts at the diodes.

approximately one million volts to appear across the field-emission

diodes. Each diode is capable of producing a 40 kJ electron beam

in a 70 ns pulse. A cross-sectional view of one line of the simulator

illustrating the geometry is shown in Figure 2.

The Casino simulator is similar to many other large-power, pulsed

devices in that lack of shot-to-shot reproducibility makes accurate

machine diagnostic essential. Since it is not possible to make direct

measurements of delivered energy and beam conditions while the

simulator is being used for nuclear weapons effects testing, electrical

diagnostics must be used to indicate if machine performance is normal

for a given shot. Without this information considerable operating

time can be wasted.

Diagnostics used to monitor the simulator functions are of two

general types, DC and transient. The DC measurements include charge

voltage and currents, water conductivity, vacuum, liquid levels, etc.

Although only the transient diagnostic measurements made during the shot

will be discussed in this paper, during simulator operation the DC

119
Figure 2. Diagram of a single, coaxial -transmi ssion line of the

Casino nuclear weapons effects simulator indicating

relevant machine geometry and locations of diagnostic

monitors discussed in text.

monitor records are frequently referred to in an attempt to explain

an anomally in the pulse waveforms in terms of the machine's pre-shot

status. This is particularly true when the transient diagnostics

indicate a severe abnormality.

Transient voltage diagnostics are located at the PFLs, the input of

the coaxial transformer and immediately in front of the vacuum diodes.

Since the voltage monitor at the input to the transformer is only used as

a timing signal for the purpose of synchronizing experiments to the output, it will not be discussed further. All these monitors are of the

capacitive divider type similar to the one illustrated in Figure 3

(manufactured by Maxwell laboratories, Inc., San Diego, CA 92123) and

produce a signal up to 10 smaller than the input waveform. Monitor

response in times of about one nanosecond are achieved by compensation

of the secondary resistive divider.

There are two types of transient current monitors located at the vacuum diode which measure the output current. cour, small, B-dot

loops are equally spaced on the diode door; outputs from all these

monitors are averaged by connecting their signal cables through power

120
INNER CONDUCTOR

Figure 3. Capacitance divider used to measure voltage on the waterfilled, coaxial transmission line. The division ratio

of this divider is on the order of 1:10$.

tees to reduce the effects of current asymmetry on the measurement.

A self-integrating Rogowski belt is positioned around the cathode

shank approximately four centimeters from the anode-cathode gap.

Comparison of the output of the two current monitors gives an indication

of any spurious current paths by-passing the anode-cathode gap.

Transient digital analyzers are used to record the waveforms and

a description of the data recovery and processing system is discussed

in Section 2. Section 3 describes the procedures used to calibrate

the voltage and current monitors, while Section 4 compares calculated

delivered energy with measurements made by total absorbing electron

beam calorimeters.

2. Data Acquisition

One of the most important parts of any data acquisition system is

its grounding method and signal cable connection procedure. For the

Casino Facility the primary grounding system provides conductance from

the radiation^ test-cel 1 s electromagnetic shielding, Marx generators

and facility data room -- where all transient simulator waveforms are

121
processed -- to earth. A second ground system is used to earth a second instrumentation room and these two grounding systems can be

connected or isolated as desired. Both grounding systems use 1/4-inch

thick, 12-inch wide strips which have 5/8-inch diameter, copper-weld,

ground rods driven into the earth at two-foot intervals. These

grounding rods are placed underground, beneath the entire simulator area with numerous surface attachment sites. Provision is made to allow

wetting of the earthing system for improvement of ground conduction,

although this has not been found to be required to maintain an adequate

signal reference.

Signals are transmitted from the diagnostic monitors to the

facility data room via low dispersion type foam-flex cables which were carefully cut to specified lengths so that timing measurements could

be made to within 0.3 ns. These cables have a solid aluminum

outer conductorswhich provide excellent noise attenuation; however, it was discovered during diagnostic calibration that the connectors

attaching to the shield room were acting as a noise source which gave

a consistent distortion of the transducer waveform. The explanation for

this phenomena was found to be corrosion of the connectors mating

threads; tightening the connector to over the specified torque was found to temporarily relieve the problem, but risked permanently

damaging the connector. A satisfactory solution was found by applying

N0AL0X (Ideal Industries, Inc., Sycamore, IL, Catalog No. 30-030) which

is used in the electrical industry to prevent connector oxidation on aluminum wiring. It is important that all cable shield are tightly

bonded to the shield room since difference in ground reference of

spacially separated locations on large high-voltage machines can generate large shield currents. Approximately 400 amps net current

have been measured on the foam flex cables which can cause sizable

voltage drops across grounded connectors unless their resistance is

very smal 1

.

Because of the noisy environment that exists when the simulator is

fired, a good quality shield room is required to record diagnostic

data. The facility data room has solid iron walls with welded seams

to limit the penetration of magnetic as well as electric fields. This

enclosure was tested and met the specified shielding requirements for

both magnetic (^35db at 1 kHz to ^90db at >_10kHz) and electric (>120db

at >_1 0kHz ) fields (Shield room manufactured by Lectro-Magnetic , Inc.,

Los Angeles, CA) . All 60Hz power is brought into the enclosure through

low pass filters that have an insertion loss of greater than lOOdb from

15kHz to 10GHz; all communication lines entering the enclosure are similarly filtered.

Provision is made to electrically isolate the facility data room. The room is placed on two inch thick insulating pads and may be disconnected from the simulator ground system. Instrumentation power can

be obtained either from a motor generator set, if electrical isolation

is required, or standard utility power. It has been found that during

normal operation satisfactory results are obtained using commercial power

with the screen room connected to the facility earthing system.

122
All pulsed data is recorded on Tektronix 7912 transient digital

analyzers with type 7A24 vertical plug-in units which gives the system

an upper frequency response limit of 400MHz, far above the approximately

three megahertz fundamental signal to be recorded. The transient

digitizers record data in its internal, 4096-word memory and this

information is numerically processed by a computer to obtain useful

waveforms. At the Casino facility data processing is performed by a PDP

11/40 using i n-house developed software.

Waveform data processing can be described as being accomplished in

five steps. Initial ly, the digitally recorded signals stored in the

791 2' s internal memory are retrieved from the individual units and

stored on disk. Each recorded data word represents a horizontal or vertical data point from the 512x512 diode matrix. Prior to storing

waveform data, software must process the raw digitized data to correct

for noise, remove points due to flaws in the diode matrix, and discriminate against multiple valued functions. Next, the units are triggered

again with no input signal for the purpose of determining a zero amplitude reference, and this information is stored in computer memory.

Then, each unit is triggered again to place an accurately timed marker

pulse on the diode matrix and this information is also stored in memory.

The purpose of the marker pulse is to time reference the data signal.

Sensor sensitivities, cable delays, and offset voltages are used in

conjunction with vertical and horizontal calibration factors to create a measured data waveform. All these operations are performed using assembly

language to speed up data processing time. Once this is completed,

the data is transformed into 32-bit real numbers so that the next step

of data manipulation using standard Fortran can be employed. After the

conversion routines are completed the raw data has been transformed to

a waveform that is the equivalent of standard oscilloscope records, but

with actual voltage and current scale factors, and time correlation.

The hard copy output, which is routinely obtained once all the

data processing is complete, contains measured waveforms of the PFL

voltage, diode voltage and current, plus calculated values of power

and energy. Selected parameters for the waveforms are also printed on the display, such as maximum voltage, time maximum voltage occurs, etc.

A typical hard copy display is shown in Figure 4. Besides the standard

output package the data may be manipulated by any waveform operation;

for example, dividing the diode voltage waveform by the current waveform

to determine the time history of diode impedance. This capability was very useful during the calibration of the diode monitors.

3. Monitor Calibration

A typical problem that arises when calibrating monitors for large,

high-voltage equipment is that often the device' s sensitivity must be

measured in-situ because of its geometrical dependence. This implies

that considerable amounts of energy must be used to supply calibration

voltage and current pulses at even modest levels. Therefore, transducers

which have voltage or current coefficients are susceptible to error

123
PLUB1 CABLE NO. 10

YMIN = -4.005E+06

YMAX = 1 524E+06

T IMEC YMIN > = 5 536E-06

TIME(YMAX> « 5 920E-06

SHOT 0300

3 0000E+Q6

PLUB1

1 0000E+06

U0LT5

-1 .0000E+06

-3 0000E+06

-5 9090E+06

USER 0960 TIME 09-04^31 DA rE 20-MAR-S1

— 2 . 3 . 30OGC 06 3 . 0C00E-06 6 . 5000G-06

Figure 4. Typical hard copy display obtained from the voltage monitor

located on the pulse forming line. Amplitude values contain

all the calibration factors and correction terms. Print

out above waveform indicates voltage maxima (both positive

and negative) and the times when they occur. since calibration levels are often 100 to 1000 times smaller than

operating levels. Furthermore, monitor frequency response must be

considered, since an increase in the calibration pulse amplitude often

is followed by slower rise-times to the point that the calibration pulse

is much slower than the event it is intended to simulate. When this is

the case, it is difficult to correct for monitor droop and hi qh frequency

response.

Because of the difficulties in producing ideal voltage and

current calibration pulses, calibration runs were made by using as many

redundant checks as possible. These checks included calculations and

independently calibrated monitors for use as cross-references. Accurate

calculation of expected values was difficult since the calibration

circuit values are not well defined; however, by careful comparison of

both wave shape and amplitude good agreement between measurement and

124
calculation was reached.

To calibrate one of the PFLs a single (2.5 y f) capacitor was connected with a gas switch to the inner conductor by a length of high

voltage cable. Since the capacitance of the PFL is only 13 nF

the maximum voltage on the PFL was approximately twice that of the

capacitor charge voltage, and the resonant frequency was controlled by

the PFL capacitance and high-voltage lead inductance. Consequently, it was possible to adjust the voltage rise-time to some extent by controlling the length and routing of the high voltage cable. This method

was used to adjust the PFL charging time to be similar to the 1.2 ys

observed during operation.

In order to make a direct measurement of the PFL voltage during

calibration a conducting rod was connected to the inner conductor and

passed through the outer via an insulating fixture. The calibration

voltage could then be measured with a standardized voltage probe and the

PFL capacitive monitor was cross-referenced to it and also compared to

the voltage predicted by calculation. A bench-calibrated Tektronix

P6015 hign-voltage probe, capable of measuring transient voltages up to

40 kV, was used as the standard. A secondary copper-sul fate

resistive divider (Manufactured by Physics International, Inc. San

Leandro, CA 94577) was used as a cross check for higher PFL voltages

(maximum output of calibrating pulse % 60kV). It should be noted that

the 40 kV limit of the standard is about 100 times smaller than the

PFL operating voltage> although reference (1) indicates that the change

in dielectric constant of water should be less than one percent at the

peak expected stress.

A typical calibration waveform is shown in Figure 5 il lustrating

the damped-resonant oscillations and the longer term decrease in PFL

voltage caused by the resistance of the water. By comparing the peak

voltages a scale factor was determined for the capacitive divider

while comparison of the capacitive monitor with the Tektronix probe

during the long term exponential decay portion of the curve gives an indication of monitor "droop". The droop time-constant was determined

by plotting the ratio of the capacitive monitor signal to that of the

high-impedance Tektronix probe. This information was then used to

correct the monitor output by using the following relationship,

, T

V (T) = Vm

(T)+J- f V. (t)dt (1)

c ' m t m

where x is the droop constant, T is the time, and V

c and V are corrected and measured voltages respectively. This correction was incorporated into the Casino software.

Voltage pulses used to calibrate the diode monitors were introduced

through the front of the simulator. Again, one of the major difficulties

to overcome is to be able to produce a large enough voltage level to

record on the capacitive voltage monitors and still have the necessary

125
I 10 /usee —| TIME

Figure 5. A drawing of the PFL calibration waveform illustrating both

the damped oscillatory portion of the resonance charging of

the water line and the long term voltage fall due to water

conductivity. The difference between the rate of fall of

the input signal and that of the monitor is due to

capacitive droop which can be determined from this waveform

and corrected numerically.

rise-time (^50 ns). In order to achieve a fast, large amplitude

calibration pulse, a compact high-voltage pulser with a low-inductance

connection was required. The connection was made by attaching the highvoltage terminal of the pulser to the cathode stalk and making a short,

coaxial, ground connection to the anode-support structure. Even with

this design, the maximum voltage signal generated was approximately 15 kV

with a 100 ns rise-time. A Tektronix P6015 was used again as reference

and was connected by radial rods that were located in the same crosssectional plane as the two, diametrical Ty-opposed capacitance monitors.

Because of the faster rise-time of diode calibration pulse the standards

were positioned as near as possible to the capacitive monitors without

changing their coupling capacitances. Initially, each of the monitors

were checked separately and their individual outputs were made equal

by use of appropriate attenuators. Then both outputs were averaged

together by means of a power-tee and calibrated for both amplitude and

droop correction. The purpose of using two voltage monitors to

measure the diode voltage is to help account for any asymetry in the

voltage wavefront.

126
The Rogowski coil, used to measure the diode current, was purchased specifically for use on the Casino diode and was supplied

with a calibration factor (manufactured by Physics International, Inc.,

San Leandro, Ca;). This factor was obtained by using a special constant

impedance (50ft) test fixture designed for the purpose of supplying a fast-rise, square waveform. These calibration factors were double-checked

at the Casino Facility by using the same calibration pulser used to test

the diode voltage monitors. The pulser was connected to the diode door

and a Pearson 301 X current monitor was mounted inside the door cavity;

a shorting rod, connected to the outer edges of the door by wide

conducting straps was passed coaxial ly through both current monitors.

The supplied calibration-factor was confirmed (to within the stated 5%

accuracy). The four B-dot current monitors were also calibrated while

the diode door was in this configuration.

After the senitivities of the current and voltage monitors were determined, the final step in the diode calibration process was started. Figure 6 shows the schematic of an elementary circuit model

of the vacuum diode. Since the voltage is measured at a location

distant from the anode-cathode gap, this measurement must be corrected

for inductive voltage drops to give the actual electron acceleration

potential. A standard procedure for making this connection is to

determine the diode inductance by shorting the anode-cathode gap

and measuring the voltage across the vacuum envelope. Once this

diode inductance is determined then the diode voltage (V^) is

related to the corrected measured voltage by

where L is the diode inductance and di/dt is the derivative of the diode

current.

Use of the digital data acquisition system greatly simplifies the

diode inductance measurement. Since the self-integrating Rogowski

coils supplied the diode short-circuit current directly and numerical

differentiation of the current signal was quite noisy, the inductance

was determined by integrating the corrected measured voltage about

the current maximum,

V

D

= V

c

- L di/dt, (2)

T (3)

127
DIODE

o—'.

VM

"DIODE ©

Figure 6. Simple lumped parameter circuit used to model vacuum diode. The measured voltage (V ) is

corrected for the voltage drop across ^Bhe

anode-cathode gap.

In general, plotting the relationship gave a smooth, but non-constant

value for the diode inductance. Values obtained for diode inductance

could be changed by adjusting the time-phasing of the current relative

to the diode voltage. The question of proper current and voltage

phasing occurs because the current is measured at a different location

from the voltage; the monitors are located approximately seven nanoseconds electromagnetic wave propagation time from each other.

The simple lumped inductance model for the diode inherently assumes an infinite energy propagation velocity, i.e, all points on the series

circuit pass the same instantaneous current. An alternative to this

simple model is to attempt to describe the diode by transmission line

analysis which becomes difficult because of the nonuniform geometry

in the diode region (reference (2)).

Use of the lumped inductance model for the ideal short-circuit

implies that (1) diode inductance is constant, (2) all measured voltage

is due to the diode inductance, and (3) no net power or energy is

delivered to the diode. Realistically it is not possible to make a

a non-resistive short circuit because the large short circuit currents

produced can cause several kilovolts to appear across even a few

milliohm connection. Therefore, the approach chosen was that of

adjusting the current and voltage phasing until an approximate constant

inductance was calculated, see Figure 7. For each short circuit shot

the time-average of the calculated inductance was made; five of these

measured inductances were then averaged to obtain a value for the

128
0 I

1 L 2.60 2.75

TIME (//sec)

Figure 7. Variation of diode inductance with current and voltage

phasing. Time shifts and associated time averaged inductances

shown are no shift and 60.0 nH (long dashes), 8 ns and 73.1 nH

(solid line), and 9 ns and 75.2 nH (short dash line). The curve determined by the eight nanosecond shift was determined

to be the nearest to a constant inductance valve during the

short circuit pulse.

diode inductance. The diode inductance was found to be approximately

70 nH with a voltage and current phasing delay of 8 ns. Proper

phasing is very important, because a shift of 1 ns changed the

calculated energy (on non short-circuit shots) about 1 kj {2% per ns).

129
4. Comparison of Calculated and Measured Energies

As an independent check of the voltage and current calibrations,

a series of experiments were conducted using total electron-beam

absorbing calorimeters. These devices, which are used to determine the

total energy of the electron beam by measuring the thermal energy

delivered to the calorimeter, are subject to losses which lead to an under estimation of the beam energy. These losses arise from: (1) back

scattered beam electrons that never enter the calorimeter, (2) radiative

cooling of the calorimeter front surface before thermal equilibrium is

reached, (3) heat loss to nearby bodies due to thermal conduction, and

(4) front-surface spall and vaporization. Methods to reduce the effect

of these mechanisms were used in setting up the energy correlation

experiments

.

The calorimeters were made from approximately .5 kg, cylindrical

blocks of AXF-5Q Poco Graphite (Poco Graphite, Inc. Decatur, Tx, 76234.)

for which the specific heat as a function of temperature was known.

Although the calorimeter was electrically connected to the grounded,

conducting drift-tube, a tortuous thermal conduction path was made to

limit energy losses. Thermal couples used to measure the temperature

rise of the carbon blocks were electrically isolated from the calorimeters

by mica washers to prevent large electric transients from entering the

high-impedance recording circuits and causing a permanent offset. A

short drift tube was used to reduce the effect of backscattered-el ectrons

.

The electrons reflected from the carbon block were radially trapped by

the magnetic field which forced them to undergo several oscillations

between the calorimeter and the diode, thus increasing the probability

of collection. In order to reduce the effect of radiative-cooling, spall

and vaporization losses, the cross-sectional area of the electron beam was expanded almost a factor of two by reducing the magnetic guide field

in the region of the calorimeter, and furthermore, the simulator was operated at half-power to reduce the delivered energy. Adjustment of

the anode-cathode gap was made for the half-power shots in order to keep

diode voltage at nominally one megavolt.

As a result of the precautions taken in making the calorimeter

measurements, good agreement was obtained between the total diode energy

measured by the carbon block and that calculated by integrating the

product of the voltage and current. Table I gives the comparison between

calorimeter measurements and the calculated diode energies and shows that

the agreement is well within the experimental error estimated as being on the order of seven percent. Unfortunately, a similar set of experiments

for full simulator power is not in as good agreement. From the

description of calorimetry sources of error previously presented, it

would be expected that the carbon block measurements should be smaller

than the calculated values. This was not the case as can be seen by

referring to Table I. A possible reason for this discrepancy appears to

be that the Rogowski current monitor is ocassionally shorted during the

power pulse by plasma bridging its electrostatic shield gap. The problem

of plasma shorting arises because it is desirable to locate the current

130
Table I. Comparison of calculated and measured diode output energy

DIODE DIODE Bl DIODE B2

Shot

No.

Cal . Energy

(kJ)

Meas

.

Energy

(kJ)

Percent

Error*

Cal

.

Energy

(kJ)

Meas

.

Energy

(kJ)

Percent

Error*

202

203

22.4

22.5

22.6

25.2

-1%

-11

-Data

1 ~ /- 4- LOSt

23.2

21.1

21 .6 +2

HALF

204 22.7 22.4 +1 23.2 24.5 -5 POWER

205 22.6 24.3 -7 23.2 22.8 +2

206-

o n 7 Simulator Malfunction No Output

208 I-Data

Lost

25.0

AVE -5%

24.1 24.0 0

AVE 0%

209

210

I-Data

Lost

35.1

39.5

39.4 = 11

33.3

32.6

42.0

41 .5

-21

-21

FULL

211 34.4 36.8 -7 30.6 41 .5 -26 POWER

212 35.2 40.8 -14 I-Data 46.6

Lost

AVE -11% AVE -23%

*Percent Error f(Ca1. Energy - Meas. Enerqy)/Meas . Enerqy)xl 00%

monitor monitor as close as possible to the diode anode-cathode gap.

Therefore the Rogowski coil is in a high-stress, high-photon fluence

environment which can lead to plasma formation and shorting of the

monitor during the latter stages of the main power pulse. As seen in

Table I, when current monitor is shorted, the uncertainty in the

delivered energy can be on the order of 25%.

5. Summary

Half-power measurements made at the Casino Facility have shown that

it is possible to get good agreement between voltage and current

diagnostics and total electron beam absorption calorimeters. This is

131
a necessary, but not sufficient indication that the transient diagnostics

are recording correct voltage and current values. Furthermore, it is

possible to observe noticeable changes of the current waveform when

computer values are suspect at full energy operation. Although this

gives some confidence on the simulator waveforms recorded it does not

imply that other possible correlation measurements, such as measured

electron spectra, would necessarily give as good agreement.

The general philosophy used in developing the Casino diagnostic

system was to incorporate wherever possible redundant measurements that

are based on different physical principles and independently calibrated.

Because it is not possible to directly relate any of these measured

quantities to accepted international standards, the policy of redundant

diagnostic monitors has been used so that at least consistency could be

shown when they are in agreement. When agreement between monitors

measuring the same parameter is poor, then at least some indication of

the measurement uncertainty is obtained.

References

(1 ) Water: A Comprehensive Treatise, Volume 1 The Physics and Physical

Chemistry of Water , Ed by Felix Franks, Plenum Press, New York,

page 262 (1972).

(2) G. A. Huttlin, "Interpretation of Remote Current and Voltage Data

in a Pulsed System," HDL-R-22900-80-1 , Harry Diamond Laboratories,

Adelphi, MD (Oct 1980).

Sponsors Acknowledgments

This work was sponsored by the Defense Nuclear Agency MIPR

No. 77-559 (1976)

.

132
INDIRECT MEASUREMENT

BY

COMPUTER SIMULATION

ROBERT A. HILL

WESTINGHOUSE ELECTRIC CORP.

BALTIMORE, MD . 21203

Enhanced instrumentation by means of computer

aided techniques is particularly useful for measurements

of fast, high energy electrical transients. An overview

presented in this paper offers a general approach to

making indirect measurements by mathematical modeling.

The illustrative examples are taken from case histories

of simulating the dynamic behavior of pulse circuits.

Various examples described herein were selected to

indicate a wide range of usage. These examples are

given without the intent of rigorous analysis or rigid

procedure. The purpose is to give discussion in

appropriate workshop style.

Mathematical modeling is particularly useful in

design and development of high power, complex systems

because of necessary time and resources that must be

committed for actual construction. The model can serve

a variety of purposes for analysis of the equipment. A

single, all purpose model would be too unweidly for

practical purposes; therefore, the model generally

consists of a series of special purpose models, or

related mathematical equations which describe a limited

function. Solution to the descriptive equations may be

determined by computer programs with suitable output

display. Some models which may be adaptable for

indirect measurements and estimating are discussed in

the following sections.

133
TYPICAL PROBLEMS

Several unique problems arise when making

measurements of large electrical transients in pulse

power systems. A few of the common difficulties are

listed below.

a. Dynamic range of instrumentation presents a limitation of measuring extreme conditions.

b. Spurious interference affecting

instrumentation, such as electromagnetic

radiation.

c. Inaccessible point of measurement due to

mechanical obstruction or high voltage

hazards. In some cases, such as loop currents

caused by stray reactance, a single point of

measurement is non-existent.

d. Masking of quantitative measurement by

extraneous sources and environmental noise.

e. Fault condition instrumentation where the

fault would cause a dangerous situation.

EXAMPLES OF SIMULATION

Several illustrations of simulation may be taken

from the line type pulser discharge circuit shown in

Figure 1. The line type pulser was chosen since it is

easily identified and offers many simple examples of

modeling. Examples presented herein are reduced from

complex problems which may exist in real situations.

These examples indicate extensions of primary

instrumentation performed by conventional means. In

fact, the exsistence of a useful model depends upon its

correlation to direct measurements.

Waveforms

A typical output waveform of current from a pulse

forming network is shown in Figure 2. This waveform is

the result of several currents which exist in the

network loop elements as a function of time during the

discharge period. In this case ii is the output

134
135
136
OUTPUT CURRENT

FIGURE 2

137
current in Rl which is readily available for direct

measurement. However, the remaining currents in various

other portions of the circuit may be of significant

interest, yet locations of measurement may be

inaccessible. If element values of the network are

known, then the remaining current waveforms may be

calculated by relating the one measured value to all

others, such as shown in Figure 3. An extension of

this procedure can be taken another step. In this case,

waveform of i4 is in an inaccessible location.

Assuming that a load fault occurance was caused by a voltage arc or other shorted conditions, the current

i4 waveform may be simulated as shown in Figure 4. In

some cases, particularly high power pulse systems, the

fault current cannot be easily measured nor is it

advisable to attempt direct measurement. This in turn

provides yet another step in indirect measurement by

simulation. If a load fault current of large magnitude

is to be prevented, then protective circuits such as a crowbar circuit must be used to divert the energy.

Other problems then arise with direct measurements such

as the dynamic range of instrumentation required and

masking effects caused by interference. Both of these

problems are present during crowbar action of high power

systems. Simulation of the circuit conditions offers a means of gaining insight into the quantitative values.

Stray Reactance

Quite a different approach of simulation may be

used to measure stray reactance of a pulse transformer.

In this case, the output pulse transformer of a line

type modulator was found to cause excessive waveform

distortion by ringing. Conventional means of directly

measuring stray leakage inductance and stray winding

capacitance failed to provide sufficient details. An

equivalent circuit as shown in Figure 5 was used for

analysis. Values of coupling between primary and

secondary windings are shown as MpS . Stray

capacitance values of the secondary windings are given

as Ci and C2 • Comparisons of actual observed

waveform and simulated waveform were made as shown in

Figure 6. Parameter values of the simulated circuit

were adjusted until agreement was reached for magnitude

and frequency of ringing. With this technique, the

effective values of stray reactance could be

138
139
140
LOAD FAULT CURRENT

FIGURE 4

141
142
143
determined. This example illustrates a means of

determining effective reactance values of an actual

circuit by comparison with known model values which were

derived by a trial and error procedure.

Magnetic Field

Consistency of scaling magnetic fields produced by

current carrying conductors in a homogeneous medium

provides a reliable means of modeling. Advantage of

this fact may be used in several cases where direct

measurements would be difficult or where the

instrumentation would introduce extraneous errors.

Figure 7 shows a calculated magnetic field plot of

two parallel, current carrying conductors. Each TIC

mark on the X-Y axes represents a unit of distance and

the surrounding lines represent equal magnetic flux

density. These lines also indicate the field gradient

change as a function of distance. In this example, it

was desired to produce a uniform field at distance Dy

from the conductors. As shown, a two conductor system

produces a very non-uniform flux gradient. Because of

dimensional limitations and interposing barriers in the

actual system, direct measurement techniques were

restricted. Field gradient at the desired points would

also introduce instrumentation errors. With the aid of

computer plotted data, it is possible to overcome some

of the restrictions. First, an acceptable model of the

conductor system was established by means of

mathematical computation. This system has a go-return

path of 8 conductors as shown in Figure 8. It

represents a multi-turn winding of 1200 amp-turns. Each

group of conductors is shown as "*" on the graph, with

spacing 1.8, 1.533, 1.266, and 1 units of distance. The

area of flux density lines shown are bounded by 300 and

500 gauss flux density lines which is the area of

interest. The next step in deriving a workable system

was the construction of a physical scale model. Scale

modeling was used to provide information about field

intensity variation and also provide a check on

computational accuracy. The scale model was made to

convenient dimensions for instrumentation. From

measured parameters, the relationships of scale model

and mathematical model could be determined.

144
145
146
Measurements were made in areas of relatively low field

gradients for correlation. After reasonable agreement

was established between the two models, computer

techniques were used to establish final design of the

conductor system. In the final system, direct

measurement of the magnetic field distribution are very

difficult to measure. However, computed estimates

provide sufficient data for practical purposes.

Dynamic behaviour of the magnetic field may be

simulated. The magnetic system described above was

designed for use in a repetitively pulsed application.

Therefore, inherent inductance is a major

consideration. Affects of inductance are implicit in

the simulation. This can be seen by comparison of

Figures 7 and 8 for flux line concentration. Although

further elaboration of this technique is beyond scope of

this paper, it can be seen that space fields may be

modeled for illustration of a particular purpose.

Data Extrapolation

Beneficial information may be generated by

extrapolation of limited measured data. This technique,

although somewhat restricted by complexity of

simulation, is useful over certain ranges for estimating

purposes. Generally, it involves a "curve fit" program

available on most computers. The system to be

extrapolated does not necessarily have to be linear, but

it does have to be consistent within the bounds of

simulation. With a limited amount of data measurement

points, certain mathematical functions can be generated

for determining relationships. However, the number of

points and the spread of data points are critical to the

degree of approximation. If, on the other hand, data

points are taken for a system having a known

mathematical relationship, the extrapolation may be made

with a fair degree of certainty. This is the case for

component temperature rise over an extended period of

operation. Temperature rise characteristics are related

as shown in Figure 9. In most cases of uniform

temperature rise, an estimate of the time taken to

arrive at a final operating temperature can be made.

Two temperature measurements taken at equally spaced

intervals of time during initial temperature changes can

be used to predict final operating temperature. With

147
148
large components having time constants in the order of

several hours, this technique provides a means of

predicting final temperature in a time period less than

one time constant.

GENERAL SUMMARY

Modeling is generally an abstract approximation of

the actual working system. Perhaps, the most important

consideration of modeling is "What and how much". Over

simplification leads to gross errors in results. An

overly complex simulation may be time consuming and

expensive beyond value of the results. Deriving a reasonable compromise is usually based on past

experience in simulation techniques. An overly

ambitious program on first attempt is very difficult to

"debug" . One approach which seems to work best is a step by step model building: Starting with only the few

essential elements as a crude approximation; then

building and adding other considerations; and finally

discarding those elements which have little significance'.

An attempt at making meaningful measurements or

even approximate estimates would seem to rely on how

complete the model represent an actual system.

Obviously, the best means of accomplishing this would be

either reduction and simplification of the system or

simulation of only limited aspects of the system.

149
CALIBRATION OF THE BLACKJACK 5 PULSE GENERATOR OUTPUT POWER*

R. Richardson, E. Chu, W. Clark

J. Shannon and M. Wilkinson

Maxwell Laboratories, Inc.

8835 Balboa Avenue

San Diego CA 92123

A discussion will be presented of the calibration techniques

routinely used to ensure precise measurement of output power from

the BLACKJACK 5 pulse generator. Because of the high voltages

(<3 MV) and high currents (^5 MA) which must be measured during

normal operation, probes with very low absolute sensitivity are

used. Calibration procedures will be described which consist of

applying voltages and currents of sufficient levels to overcome

the low sensitivity and provide good signal-to-noise ratios. The

calibration pulsers are also designed with outputs having dominant

frequency components close to the measure signals. Cavity type

§ current monitors are calibrated in in situ and corrected for wall

effects. These monitors are also checked against B* loops. E type diode voltage monitors are also calibrated in situ by measuring

the probe capacitance and by direct comparison with a voltage

divider. Careful attention is paid to the frequency response of

passive RC integrators used in the calibrations and final measurements. All cable lengths are trimmed to ensure precise phasing of

the signals, and FFT techniques are used to correct for cable

attenuation

.

*Work supported by the Defense Nuclear Agency.

1. Introduction and Background

This paper presents the results of a set of experiments

designed to measure the output power on the BLACKJACK 5 generator

at Maxwell Laboratories. BLACKJACK 5 (shown in Figure 1) is a water insulated transmission line structure with four time com- pression stages. The first stage (transfer capacitor) is charged

with a Marx generator and sequential closing of water switches

150
transfers the energy in successively shorter times to the output

(diode). The output (diode) is a water-vacuum insulator and vacuum

transmission line which converges the energy to the load. The

machine is used to produce dense plasmas by imploding wire arrays,

foils and gas jets. The nominal output parameters are:

Voltage 3 MV

Current 5 MA

Power into the diode 10 TW

Energy into the diode 500 kJ

Pulse width 50 ns

Other pulse widths can be achieved by switch variations but the

50 ns mode is the usual one.

INTERMEDIATE INTERMEDIATE TRANSFER PULSE-FORMING PULSE-FORMING

TRANSFER CAPACITOR CAPACITOR LINE LINE TRANSFORMER DIODE iQjrin

Figure 1. BLACKJACK 5 pulseline —

cross section of structure,

An accurate determination of the power is important for two reasons. First, some of the plasma parameters (such as radiated

energy) appear to depend strongly on the diode input power. For

this reason, an accurate power measurement helps verify scaling

laws which are used to determine future machine requirements.

Second, there are certain design goals (such as 10 TW). It is

necessary to know the power to sufficient accuracy to demonstrate

that the goals of the program have been achieved.

In January 1980, the opinion was voiced that the Maxwell

BLACKJACK 5 system did not achieve the 10 TW that had been advertised.

Because of this we began an intensive review of all the diagnostics

to determine the actual power of the system into the diode load.

The effort included the calibration of the pertinent data channels

and careful scrutiny of all effects (such as risetimes) that might

influence the power measurements. The result of this effort is

presented below.

151
2. Factors Affecting Diode Power Measurement

Along with the usual calibration accuracy, there are additional

factors that have to be considered in determining the power into the

diode. Examples of these are the relative phasing on the current

and voltage waveforms and such effects as high and low frequency

degradation. Since the present diode voltage has a large inductive

component, the V and I waveforms are out of phase. This exaggerates the effect of phasing erors and frequency cutoffs.

2.1 Cal ibration Accuracy

The accuracy with which the voltage and current probes can be

calibrated is the most critical factor in the power measurements.

It is clear that to get a ±10_percent power measurement a ±5 percent

calibration must be done, this calibration is discussed below and

in other papers presented at this conference [1] .

2.2 Voltage and Current Phasing

Because the diode is largely an inductive load during the

early part of the pulse, dl/dt is near maximum at peak voltage.

This means that an error in the phase (relative timing) of the

voltage and current waveforms has maximum the effect. For an inductive load, the effect of a time shift in the peak power is

given by

t-l+f (1)

P 'o

where P/P is the normalized power AT is the time shift of the

voltage with respect to the current and T0 is the width of the

power pulse (FWHM). The effect of a time shift on the BLACKJACK 5 waveforms is shown in Figure 2.

At first it appears that the only issue here is the correct

trimming of cable lengths to the desired accuracy. There are,

however, some subtleties that must be considered. In this machine,

as well as others, the voltage measurement is made by a capacitive

probe upstream of the diode insulator ^2.5 ns away from the current

probe which is in the vacuum. This raises the question as to whether

voltage waveform should be delayed by this time difference. This

question can be answered by considering the circuit of Figure 3.

This shows the relation of the diode voltage probe with respect to

the current probe in the BLACKJACK 5 generator.

152
T

H = POWER PULSEWIDTH (FWHM) = 50 ns 1,36

1.281-

1,20

| 1,12

o

| 1.04 « 1,00

UJ- 0,96

i =cSI

en§ 0.88

0,80h

0.72

0,64

1 +

1 1 1 1 -15 -10 -5 0 5 10 VOLTAGE TIME-SHIFT (ns) RELATIVE TO TRUE POSITION

15

Figure 2. Plot showing the sensitivity of the power to an error in voltage and current position.

WATER LINE

VOLTAGE

PROBE POSITION

CURRENT PROBE POSITION

V

VOLTAGE, CURRENT AT VOLTAGE PROBE POSITION,

VOLTAGE AFTER TIME DELAY T,

VOLTAGE, CURRENT IN DIODE.

Z

0 , T0 , L0

= IMPEDANCE, ELECTRICAL LENGTH AND INDUCTANCE OF TRANSMISSION LINE BETWEEN DIODE AND VOLTAGE PROBE.

Zg = 0,6f2, Tg = 2,5 NS, Lq = 1.5 nH

Figure 3. Relation of voltage and current probes

153
This circuit can be analyzed in the frequency domain. For times

long compared to characteristic electrical lengths (AT, T0 ) the

solution can be expanded for small values of frequency and the

coefficients of various derivatives can be identified. The voltage

at the oscilloscope V* is then related to the diode voltage

by

dI

o dV

D V* ~ V n + Z T —£ - AT w ~ D o o dt dt

d

2

I n AT

2 + T

2 d2 V n

Z T

3

d

3

I n

_ Z T AT -4 + ° + 1 +

0 0 dt

2 2 dt

2 6 dt

3

(2)

It can be seen that adding a time delay AT in the voltage channel

to account for the probe position adds terms proportional to

dVn/dt and I p/dt^ . The term added by the transmission line is

proportional to dln/dt in this limit and cannot be cancelled by

the dVp/dt term. For this reason, it is better to make

AT = 0 , in which case there is a first derivative correction term

in the current and the next corrections are second derivative in

voltage or higher. In this case, the voltage at the probe is

related to the voltage at the diode by

dl n T

2

d

2

V n

Z T

3

d

3

I n

where we have put L 0

= Z0 T0

. The power defined by the probe

voltage and diode current is then

= V

D

I D + al ItL„V) + •• (4)

From this we see that the power defined in this fashion is the true

diode power plus the time derivative of the magnetic energy stored

in the transmission line between the probe and the diode. This is

all consistant if the diode is considered to be the volume downstream

of the voltage probe. Typically, the second term in Eq. 4 is

^5 percent of the diode power. Hence, the power measured in this

fashion is that which flows past the voltage probe.

154
2.3 Frequency Roll-Offs

Another factor which affects the power measurement is the

frequency response of the diagnostics used in measuring the voltage

and current. These effects can be characterized in the high and

low frequency limits by frequency cutoffs. There are three separate

effects, one at low frequency and two at high frequency.

The low frequency roll -off is caused by finite integration

time of capacitive probes or Rogowski coils or the finite time con- stant of a passive integrator used with differential probes. In the

time domain these manifest themselves as a droop in the later parts

of the waveform. In the frequency domain the low frequency roll -off

is described by a transfer function

3 (k) \M =

l ' r (5)

1 +

where f|_ is the cutoff frequency. For a passive integrator, f. is given by

f

L = 2^RC (6)

where RC is the time constant of the integrator. A typical

value of f|_ is ^3 x 104 Hz (RC = 5 us). The degradation of the

peak power due to the low frequency roll -off is shown in Figure 4 vs the cutoff frequency.

The first of the high frequency effects is a simple 1/f

type degradation at high frequencies. This could be due to stray

capacitance or inductance at the probe or in the passive integrator.

In the frequency domain it is described by the transfer function

1 +

(7)

where f^ is the cutoff frequency.

In the time domain this would appear as a risetime degradation.

The effect of the high frequency cutoff on the power is shown in

Figure 4. A typical value of fu is ^108 Hz.

155
POWER PULSE WIDTH = 50 ns

LOW FREQUENCY HIGH FREQUENCY

' < ! i—l 1 1 i_l Lj L_l

10

4 10

5 106 10

7 10

8 10

9 10

CUT-OFF FREQUENCY (Hz)

Figure 4. The effect on the BLACKJACK 5 power measurement

of high and low frequency degradation.

The second type of high frequency attenuation is that due to

cable losses. In the frequency domain, the transfer function has

the form

where f is given by

Here, Z0 is the cable impedance, x the length of the cable and a a constant describing the high frequency attenuation. Typical

values of fc are 109 -10^0 Hz. These are very large but since the

frequency appears as VT~, the effect persists at frequencies in

the range of interest. The effect of the cable cutoff on the power

is shown in Figure 4.

156
All of the above frequency roll-offs can be compensated . In our case, the compensation is done in the computer software. This

will be discussed at more length below.

3. Diagnostic Calibration

In the BLACKJACK 5 diode, the probes used to infer the power

measurement are a capacitive voltage probe in the water near of the

diode interface and a cavity type current probe on the vacuum side

of the insulator. These are shown schematically in Figure 5. The

voltage is measured at three positions and the current at six

positions. This was done to average out the non-transmission line

modes of the water PFL and irregularities in the current waveform

due to losses. The individual probes are connected together and a single cable is run to the screen room for each diagnostic.

PASSIVE

INTEGRATOR "QETERMINATION

RESISTOR

E PROBE

(THREE PLACES)

TORECORDER

^VACUUM

LOAD

B CAVITY PROBE

(SIX PLACES)

Figure 5. BLACKJACK 5 diode diagnostics.

3.1 Voltage Probe

The diode voltage probe basically consists of a small capacitance with respect to the pulseline central conductor, connected

via a 50 Q cable to a passive integrator. Two methods were used to

calibrate this probe. The first was a direct comparison with the

output of a 1000:1 resistive voltage divider obtained from Los

Alamos Scientific Laboratory. The divider had previously been

157
checked out and found to give nominal 1000:1 division over a wide

band of frequencies centered on ^3 MHz. A fast-ringing discharge

of approximately that frequency was injected at the diode axis

using a 20 nF, VI00 nH, 30 kV capacitor bank. The resistive divider

(^50 kfi impedance) was connected at the diode periphery as near as possible to the voltage probe location. The outputs of the two

probes were compared directly using Tektronix 485 oscilloscopes.

The second method of voltage probe calibration, dubbed the

"capacitance-current" (CC) method involves measuring the coupling

capacitance from the high voltage terminal to the E-field probe.

It can be shown [1] that ratio of the probe current to the input

current is proportional to the ratio of the probe capacitance to

the pulse! ine capacitance. Thus, the essence of the capacitancecurrent method is to determine precisely this ratio since the

pulseline capacitance can be accurately measured. This is accom- plished by monitoring the charging current injected by a capacitor

into the cental conductor, which flows to ground largely as displacement current through the pulseline capacitance. A portion

of this displacement current is intercepted by the voltage probe and

is responsible for the probe signal. Comparison of the probe signal

with the signal provided by a Pearson current transformer monitoring

the charging current gives the desired capacitance ration. This is

discussed in detail in reference [1] .

The two methods of voltage probe calibration were in excellent

agreement with each other, giving estimates of probe sensitivity

which differed by two percent. The average of the two methods was

the sensitivity used in power calculations.

3.2 Current Probe

The injected current used for the CC voltage calibration was also used in calibration of the current monitor^ The latter consists

of a precisely machined circular groove (0.20 cm area) just inside

the insulator. A voltage proportional to dl/dt develops across

the groove inductance and is integrated with a passive integrator to

produce a signal proportional to I . The calibration used the

^3 MHz, 20 nF, M00 nH pulser described above and consisted of a direct comparison between the integrated output of the diode current

monitors and the signal from the Pearson transformer, which monitored

the injected current.

Since the wall material of the diode current monitors (aluminum)

has nonzero resistivity, a small frequency-dependent correction must

be made in order to eliminate a contribution to the unintegrated

signal due to finite wall resistance [2] . Fortunately, this problem

is well understood and since data acquisition is via Tektronix 7912

transient digitizers, the correction may be made using computer

software, as will be discussed below.

158
3.3 Synchronization

Synchronization of the current and voltage was accomplished by

precise measurement of all cable lengths using time-domain reflectometry. The instrument used was the Tektronix 1502 TDR. A ^10 ns wide pulse is generated in the screen room during each shot and is

added to all diode waveforms. This pulse, triggered by the output

switch of the pulseline, provides a fiducial timing mark for purposes

of synchronization.

4. Compensation Procedures

Because the data is acquired on transient digitizers, the

compensation for the various effects mentioned above is easily done.

The software required is relatively straightforward and has been

checked against test waveforms to determine its accuracy.

4.1 High and Low Frequency Compensation

In our case, the high and low frequency cutoffs arise from

the same circuit element, the passive integrators used in both the

current and voltage measurement. Since they are so far apart in

frequency (3 x lCr vs 10° Hz), they can be treated as two independent

effects. The correction routine has the form,

VTrue = V + TML/

Vdt + 2^ f (10)

typically. The two frequencies f|_ and fH are determined from a transfer function versus frequency plot of the passive integrator.

The second term in Eq. 10 is due to the low frequency cutoff and

usually makes the largest correction. The third term, due to the

high frequency roll-off, is usually small. However, it makes the

waveform appear to have a time shift. This can cause an apparent

discrepancy between current/voltage relation determined from the

measured cable lengths and the actual position (a few ns difference).

For this reason, this correction must be done.

4.2 Cable Correction

A method of compensating for the coaxial cable attenuation/

dispersion has been developed [3] that works well in our parameter

range. The correction has the form

159
V

7rue " V + /

K <* - T > 3?

* («)

where K(t) is given by

K(t) = §S — t > o . (12)

Here, Z0

is the impedance of the cable, x is the length of the

cable and a is constant of the particular cable and is a measure

of the high frequency attenuation. The constant a can be determined from published data, but it is more convenient to adjust a by

correcting a test waveform as described in reference [3] .

4.3 Cavity Monitor Wall Resistivity Correction

As seen in reference [2] , the finite resistance of the cavity

current monitor leads to an error in the latter parts of the pulse.

It was shown in the reference that an approximate compensation could

be done by adjusting the integrator decay time. However, an exact

compensation can be done with the computer software.

Following the analysis in reference [2] , the impulse response

of a cavity current monitor in Laplace transform space is

H(s) = 1 + 7= (13)

c

where T" c is a cavity time constant given by,

T

c

- v ft)

2

< 14 >

Here, a the conductivity, A the area of the cavity and D is

the perimeter of the cavity. For

? the current monitor in BLACKJACK 5,

1/g = 4 x 10-6 Q-cm , A = 0.2 cm , D = 2.85 cm , giving

Tc

= 15.5 ys . The second term in Eq. 13 leads to an over estimate

of the current in late time. For the present monitor the error is

^5 percent at peak current.

160
Fortunately, this impulse response has an inverse which allows

us to write

True 1+ T

c

s (15)

In the time domain, this becomes

I True

(t) = I(t) - j I(t) G(t - t) dT (16)

where

G(t) - exp (17)

This removes the wall resistivity error from the current pulse. It

should be noted in passing that cavity monitors are only practical

with high conductivity metals (copper, aluminum, . . .)• In metals

like stainless steel, the cavity time constant becomes comparable

to the pulse and large distortions occur.

5. Output Power Measurements

Power measurements have been made with short circuits and plasma

loads. The voltage, current and power are shown in Figure 6 for a titanium plasma load. The pulse shape shown is typical of the plasma

load shots. In this mode, the power pulse is between 50-60 ns wide.

At the highest level of operation, 10-11 TW are generated at the

diode envelope and 500-600 kJ are delivered to the diode.

161
12.0

SHOT NUMBER 892 CURRENT (MA) VOLTAGE (MV) POWER (TW)

DATA TRUNCATED

AT THIS POINT

0 80,0 160,0 240,0 320.0 400,0 480.0 560,0

TIME (ns)

Figure 6. BLACKJACK 5 waveforms with titanium

plasma load.

Two checks have been made on the consistency of the diagnostics

One check in the linearity of the diagnostics is to calculate JVdt

and compare it to the current I . For a short circuit, these

should be identical in shape with the proportionality constant

being the inductance. These quantities are shown in Figure 7 for

a short circuit shot. The fit is very good during the main pulse.

During the prepulse, the resolution of the transient digitizer

makes the fit less accurate. After 280 ns in the figure, the

input voltage waveform is truncated by the end of the trace so the waveform is not realistic after this time.

The output power scaling is shown in Figure 8 as a function

of Marx generator charging voltage. The straight line shown is

proportional to . This would be the case if the machine electrical parameters stay the same at different levels and if there

are no nonl ineari ties in the diagnostics. The probable error on the power measurement is about ±11 percent due to the estimated

accuracy of the calibration. The highest power achieved was 11 TW,

confirming that BLACKJACK 5 can achieve powers in excess of 10 TW.

162
10,0

5,0 -

2,5

SLOPE = 2

1,0

10

J L

15 20 25 30 40 50 60 70 80 90 100

GENERATOR CHARGE VOLTAGE (kV)

Figure 8. Scaling of BLACKJACK 5 output power

with Marx charge voltage.

163
6. References

1 M. Wilkinson, E. Chu, "Calibration of Capacitive Voltage Probes

in Water-Dielectric, High Power Pulse Generators", NBS Workshop

on Measurement of Electrical Quantities in Pulse Power

Systems, 1981

.

2 J. Shannon, E. Chu, R. Richardson, M. Wilkinson and

C. Trivel piece, "Cavity Current Monitors", NBS Workshop on Measurement of Electrical Quantities in Pulse Power Systems,

1981.

3 C. Trivelpiece, R. Richardson, J. Shannon, J. B. Smith,

"Digital Correction of Cable Attenuation Losses", NBS Workshop

on Measurement of Electrical Quantities in Pulse Power Systems,

1981

.

164
Discussion Session 3

Gene Nolting (Naval Surface Weapons Center). Now we come to the

group therapy part of the session in which everybody gets to participate

Are there any questions?

Rill Boyer (Sandia). I have a question for John Shannon. I notice

that you use hardware integrators. It seems if you were using waveform

digitizers that you could software integrate and eliminate the

corrections for low frequency droop.

John Shannon (Maxwell Laboratories). That is correct. Rut, you

have to think how historically we got into this. We are relatively

new at the transient digitizer business and the initial problems

that we had with them was that because of the base line shift, we

used to have errors in the integrations. The feeling at that time

was since we had these integrators with reasonably good frequency

domain properties, we would use them. Now, I think that we would

replace them.

Neville Harris (Ion Physics). You were referring to the phase shift

between the current and the voltage caused by the voltage monitor

being some distance from the diode. Well, that is fair enough,

providing that you don't have any reflected waves from the diode. If

the diode is mismatched from the line, then the reflected wave from

the diode is "after" in time rather than "forward" in time. If you try

to extend the machine by working into a mismatched diode, you can run into very severe problems.

Shannon . These diodes are always a mismatch because they are essentially inductances. The analysis that I gave to you in the

talk, where I make the frequency expansion, includes that effect.

That was derived by doing the transmission line theory correctly,

expanding in small powers of s and identifying the various powers of

the derivatives. As far as I can see when you are talking about a few nanosecond transit times, it doesn't make any difference. It is

not as if you are looking for a wave going past and then waiting

sometime later and getting a bounce. That would be too high a frequency effect for that analysis to deal with. Our monitors are close enough to the diodes so we feel 2 nanoseconds doesn't cause any

problems on that line.

Nol ting . I would like to make one comment. We use the foam flex

cable referred to by John Shannon. The cable is connected to the

machine at various points and then comes back to a screen room where

all of the cables are connected to approximately the same point on the screen room. During the calibration exercises that we went

through we found a distortion of the waveform, by comparing our calibration pulse measured with the Tektronix probe to the calibration

pulse as witnessed by the capacitive dividers. Upon investigation,

we discovered that there were sizeable currents flowing in the solid

165
aluminum shield of the foam flex which generated noise at the cable

connector and when we tightened down on the connection to the screen room the phase shift went away. Unfortunately after a period of

time, the problem would return and one never knew in any particular

portion of the data cycle when this phase shift would reappear because

of the oxidation of the aluminum connector. We solved the problem by

using a substance called Noalox. I think that the use of Noalox is a standard operating procedure for most electricians who work with

aluminum wiring. Since that time, we have bypassed the problem. I don't know if you ever saw anything like that or not, John, but I thought I would bring that out as something you might want to look for.

Ray Leeper (Sandia). Frank (Young), when you are measuring the

deuteron voltage by looking at the neutron yield in the forward

direction and the deuteron current by looking at 90°, I think you are assuming that the beam is normal to the target. I wondered how

sensitive is the measurement to this assumed alignment.

Frank Young (Naval Research Laboratory). For the calculations in

figure / of my talk, it is assumed that the deuterons are emitted

from a flat anode foil (flat geometry) and proceed in straight lines

toward a CD2 target which is located inside a hollow cylindrical

cathode. In a focusing geometry (e.g., a diode with a curved anode),

the ions are projected normally from the anode into the region within

a hollow cathode. For a target placed at this location, the ions are incident on target with a broad spread of angles. In fact, the

half angle in our experiments is 35°, and if the detector is located

at 15° to the diode axis, angles between the deuteron and neutron

directions range from 0° to 50°. Results in focusing geometry are presented in figure 1 of this section which are analogous to those

in figure 7 of my paper in a flat geometry. If you look in the

forward direction (15°), the energy resolution or time resolution is

somewhat larger than it is for the flat geometry case (compare bottom

of figures 7b and lb) and at 85° it is much larger (compare bottom

of figures 7c and lc). But even for this focusing geometry, the

time resolutions are small compared to the total widths of the neutron

signals. The widths of the neutron signals are dominated by the

duration of the deuteron current where it overlaps the voltage pulse

(see figure la)

.

John Shipman (Sachs Freeman/Naval Research Laboratory). I would like

to ask John Shannon about the probes he showed feeding a junction.

He put a resistor from the outer conductor to the case of his machine

to absorb the reflection. Somehow, I feel like I have asked a sucker

question, but there must be something tricky here.

Shannon . At each probe there is a 50 ohm termination. It is in

parallel with the current sensors since they are voltage probes and

it is in series with the E-field sensors. So the reflection going

back from this junction goes back into the cable and gets absorbed.

If a signal from one of the probes comes down the cable, it sees this

166
9

TIME (nsec)

TIME (nsec) TIME (nsec)

Figure 1. Comparison of neutron TOF traces obtained using a focusing diode geometry with calculations using the energy

and current traces displayed in (a). The measured TOF traces

(solid curves) are compared with calculated yields (open points)

for detectors at (b) 1.91 m and 15° and at (c) 2.36 m and 85°.

The results of calculations for a mono-energetic deuteron pulse

of 2-ns duration are shown at the bottom of (b) and (c).

167
mismatch in impedance. Part of it gets reflected back the same cable

and part of it gets transmitted down each of the other five or six

cables. When these reflected currents get back to the diode, they

get absorbed by the resistor back there. So there is not a multiplicity

of reflections that give you sort of a noisy waveform.

Shi pman . It looks like the resistor would have to be the V probe to

the outer conductor to be determinant.

Shannon . OK. The problem is an error in the viewgraph. The resistor

should be from the probe end to ground. (There was an error in

figure 5.)

Marco Pi Capua (Physics International). I have a question for

Frank Young. I remember vaguely from my old days of electron beam

focusing that one could use a gamma reaction with a very sharp

threshold for calibration of voltage monitors. It was a 1.4 MeV

gamma that had a very sharp threshold and you could actually determine

whether you had at least 1.4 megavolts or not. You could check the

peak voltage against this reaction. I was wondering whether you know

of any reaction with a threshold below 2 MV.

Young . The lowest gamma-ray induced threshold I can think of is the

photo-disintegration fo the deuteron at 2.2 MeV.

Leeper . There 4s a y-n reaction on Be at 1.7 MeV and a y-n

reaction on deuterium at 2.2 MeV. Then, all the rest of the photonuclear

reactions have thresholds above about 6 MeV.

Dan Schweickart (Wright-Aeronautical Lab). I have two questionsg.

First, for Gene Nolting concerning the attenuation for your high

voltage dividers. If I understand you correctly, you are doing all

of your attenuation for your high voltage dividers quite close to the

output of the generator.

Nolting . That is true. The Physics International probes, which are resistive dividers, are divided down to about 200 volts for about a 4 megavolt signal and then we attenuate those signals further inside

the screen room. The capacitive dividers provided by Maxwell, however,

have low voltage outputs and vary from 1 to 10 volts depending upon

the particular divider. We typically read them with Tektronix 7A24

scope preamplifier which means that we have to further attenuate down

to one volt.

Schweickart . Have you done any work in trying to couple a larger

signal voltage into the screen room and then attenuating inside? You

may find that this will help to eliminate some of your problems in

the phase shift that you were talking about.

Nolting . I have mentioned two phase shifts associated with the

diagnostic waveforms which are totally unrelated. First, the problem

168
which I spoke of in this discussion session which was due to a poor

ground connection. Second, there was the phase shift that I discussed

in the presentation which was due to the relative phasing of voltage

and current. I believe that you are talking about the connector

phase shift problem. Certainly, in general, I would say that one would be wise to bring a larger signal into the screen room and then

attenuate it there. However, we have found that the signals have

not been very noisy even with the low voltage levels at which we operate. I think that that is attributable to the foam flex cable

itself.

Schweickart . One last question concerning the use of the Tektronix

probe. I assume that if you are using the standard high voltage

probe, your screen room facility is relatively close to the measuring

point. Is that correct?

Nol ti ng . Actually not. The Tektronix probe was measured on an osci 1 loscope that was located next to the machine and the signal that was sent into the screen was read on a Tektronix 7912 transient

digitizer. These two waveforms were then compared both in waveshape

and in amplitude, but the signal from the Tektronix probe was not

transmitted further than its standard 10 ft cable. It would have

been nice to run them together and do a point by point analysis with

the 7912, but that was not feasible.

Jack Katzenstein (Physics International). I was wondering if you

couldn't eliminate some of the errors in your beam calorimetry

measurement by putting the carbon target inside of a "Hohlraum."

No! ti ng . I guess that I don't understand your term "Hohlraum."

Katzenstein . What you would have would be a hollow space, with a small hole as an inlet which may be one problem because the beam is

pretty big.

No! ti ng . Our beam diameter, without expansion, is only about 2 inches.

Katzenstei

. OK. Well then, make the enclosure 6 to 12 inches in

diameter and put the target inside of that. Make it of low heat

capacity, such that when you measure the temperature rise of the

whole thing, you essentially first measure the target absorption

plus any energy losses from radiation, or from spalation, or back

scattered electrons which get included in the sum.

Nol ti ng . Well, that is an interesting suggestion. As you recall, I discussed the calorimeter loss mechanisms which I expected would

limit accuracy and how we tried to correct for these. In low energy

operating mode, we did get good agreement with the techniques we

used. At full energy, the error. was in the opposite direction

anticipated, that is, our calculated values were lower than our

169
measured values. So I don't think your suggestion would help in that

regard. Our errors appear to be due primarily to current cutoff.

The Rogowski coils are in a shielded enclosure with a cavity opening

to allow the flux to enter. These are located about 3 centimeters

from the actual AK gap. I believe that the gap is being shorted

after a certain period of time and therefore the current hangs up.

Shannon . I would like to make a comment on the differential probe

vs. the integrating probe. We used differential probes so we could

beat this noise problem. You can get into trouble with integrating

probes because you are handling low voltage in a rather noisy

environment. So we always went the opposite route and brought hundreds

of volts into the screen room allowing the integrator to take it down

to the volt level. This is great, except you have to be careful that

in some cases you can start mixing slow signals with very fast signals.

On breakdown, for example, you can generate kilovolt spikes that

scopes and other instruments don't like very well. If you are not

very careful, your differential measurements can blow things out.

But they do have the advantage that they are very much noise immune

to problems like leaky connectors.

Jim Fockler (Physics International). A question for Gene Nolting.

You attributed the fact that your Rogowski coil had a funny waveshape

to clamping at the end due to flashing the gap. How were you able to

determine that was the clamping mechanism as opposed to the feed

flashing somewhere which would also give you constant current around

that monitor?

Nolti ng . To tell the truth, I cannot distinguish between the two

strictly from the waveshapes. However, if the feed were flashing

over, and shorting the diode at some point in time, that should

short the energy that you are delivering into the diode from the

transmission line. Recall that the energy measured by the carbon

block was 30 percent or so higher than what we had calculated and

there is not enough energy inductively stored in the diode when it

shorts to account for this large of a difference.

Jim Sarjeant (Los Alamos). Gene (Nolting), as I understood it a moment or two ago, you were using one of the Tektronix probes as one

of your measurement probes. We have had the experience of working

with the P6015 probe for several years and have developed a version

of it that is capable of operation to 60 kV. Our system utilizes a hybrid amplifier that can drive up to 1 00 m of cable so that you no longer have to worry about having the device close to your experiment.

These modified devices now have a reproducibility of about half a percent and a long-term drift characteristic of 1 to 2%. The only

problem with them is the fact that you have a line amplifier device

that requires battery power and copper box screening. We have executed

a number of i ntercomparisons between these probes and the probes that

John Power has developed and found that the Tektronix probe performs

170
(a)

Figure 2. (a) Modified high voltage probe.

(b) Circuit board and assembled line driver.

171
best used nose down (see figure 2a).

The 6015 line driver (see figure 2b) developed at the Los Alamos

National Laboratory is specifically designed to be used in conjunction

with the Tektronix P6015 high-voltage probe. Due to the short length

of the P6015 probe cable and its termination/compensation box, it is

necessary to have the diagnostics physically close to the point of

measurement. Using the 6015 line driver makes possible the remote

location of diagnostics with minimum degradation of response while

retaining simultaneous LDC mensuration capability. Typical risetimes

of the 6015 line driver into a 50- fiload (measured at the output)

are 4 ns. When driving a long length of coaxial cable, say 70 m,

the risetime at the cable end measured typically 8 ns.

Connections to the 6015 line driver are straightforward, but for

optimum response characteristics a few items should be mentioned. A

special printed circuit board was designed to minimize stray

capacitance, component lead length, and thus ringing in the LH0063.

The circuit board fits inside a Model 2411 Pomona Box and solders

directly to the Type N connectors mounted on the box. Type N connectors

are used for best pulse performance. A schematic (see fig. 3a) and

parts lists (3b) are attached. Compensation of the P6015 probe and line

driver is nontrivial. With a fast risetime signal connected to the

P6015 probe, the Tektronix instructions for compensation can be

followed except that C2, the 4-15-pF variable capacitor, adjusts only

the droop of the pulse, not the flatness. The compensator box on the

P6015 probe adjusts the risetime and the flatness of the calibration

pul se.

The maximum output of the 6015 line driver is 10 V. An overall

attenuation factor of the P6015 probe and 6015 line driver will be in

the range of 10,000 - 12,000, and the precise value is determined

when compensation is completed.

Ron Mc Knight (NBS). How much does the modification cost?

Sarjeant . The modifications to the Tektronix P6015 probe (see

figure 2a) are straightforward and involve removing the internal

plastic ring and adding a 3/8-inch radius corona ring around the end

of the metal shield at the probe nose. The probe is used nose down,

placing the high voltage on the bottom, and the grading ring is placed

around this nose. The company will make a system with a cable that

is 24 ft long. The probe output is sent into a very high speed hybrid

operational amplifier, giving about a 10-V output signal with a dynamic range of about 500-1. If anyone is really interested in it, we would be pleased to help.

McKnight . I have a general question to the audience. In assembling

the abstracts and sending them out for evaluation and putting the

agenda book together, I was struck by the lack of papers in this

172
INPUT R3 LOAD OUTPUT

Parts List

CI 3 pfd 50 V ceramic

C2 4-15 pfd 50 V ceramic variable

C3,C4 4.7 ufd 35 V tantalum

Rl 909 kohm carbon

R2 51 ohm carbon

R3 97 kohm carbon

IC National Semiconductor LH0063K

Special ground plane printed circuit board is recommended

to obtain best performance.

Figure 3. Schematic and parts list for high voltage probe

1 ine dri ver.

173
general area of power and energy measurements. I was curious why.

Is it a matter of because no one is really interested or because it

is felt to be trivial or because it is so hard? Is it generally

felt that it is good enough to measure V&I somewhere and that will

tell you the-power output? Or is it that you take what you can get?

Pi Capua . I think that, in a sense, the difficulties lie within some

of the problems that Frank Young mentioned. In these superpower

generators, the energy densities are so large that the ability to

obtain a total energy measurement before the detector disintegrates

is very very limited. Calorimeters blow up, and activated materials

get lodged in every corner of the apparatus.

Local power density measurements become difficult because of the

magnitude of the electric and magnetic fields associated with high

power densities. Voltages except for particle energy measurements

cannot be made directly and magnetic field measurements are exceedingly

difficult. In addition, non-symmetric power flow limits the accuracy

of extrapolation from local to total measurements.

174
Session C - Current Measurements

ROGOWSKI COILS, FLUXMETERS, AND RESISTORS

FOR PULSED CURRENT MEASUREMENTS

Marco S. Di Capua

Physics International Company

2700 Merced Street

San Leandro, CA 94577

This paper discusses the frequency response of current sensors for

pulsed current measurements and sources of error measurement in these

sensors. We examine the operating principles of Rogowski coils and

fluxmeters from a lumped parameter as well as a transmission line viewpoint. This second, less conventional approach, is necessary when transit times in the sensor are comparable to the time resolution of the

measuring circuit. Our discussion of current viewing resistors (shunts)

emphasizes the transient behavior associated with diffusion of magnetic

field in the sensor, and the limitations on the sensor imposed by Joule

heating.

The choice of measurement method, based on the above analysis, will

depend on the frequency, amplitude, distribution of current, as well as

on the sensitivity and bandwidth of the recording device.

1. Introduction

Fluxmeters [1], Rogowski coils [2], current transformers [3], and

current viewing resistors (CVR) [4] have been used extensively in the

past [5] to measure currents in pulsed power systems and surge currents

in electric distribution systems. The purpose of this paper is to:

(1) describe these devices; (2) summarize their theory of operation;

(3) explore sources of measurement error; (4) indicate design features

which minimize error; (5) describe the methods of calibration; and

(6) display some representative measurements. The limited space

allotted to this task requires that I rely on published literature for

many of the detailed derivations.

2. Field Coupled Current Sensors

A field coupled current sensor utilizes the magnitude of either

currents or voltages induced in the sensor as a measurement of the pri175
mary current. Depending on the geometry of the windings, a field

coupled sensor may be a fluxmeter, a current transformer, or a Rogowski

coil

A fluxmeter is simply a single or multiple turn loop. The flux

linked by the loop is <{> = N / B • dA, where N is the number of turns.

When the field is homogenous over a scale comparable to the dimensions

of the loop, then the flux linked by the loop is a measure of the local

magnetic field normal to the loop. To measure a current with such a loop we require either ( 1 ) an a priori knowledge of the current density

distribution or, (2) a distribution for calibration, which is identical

to the distribution being measured. Despite these limitations, the

fluxmeter is still a very useful device for pulsed power measurements.

A high permeability core in a current transformer couples the

magnetic flux of the primary current I into the current sensing

(secondary) windings. The electrical measurement literature treats such

transformers thoroughly [6].

The Rogowski coil is a current transformer of deceptively simple

geometry. The shape of the curve, c (formed by the coil) and the

current distribution within it do not affect measurement when simple

conditions are met. The flux linked by the coil is

$ = / / n(B . dA) <U (1 ) c A

-»• where n i^s the number of turns per unit length, B is the local magnetic

field, dA is the area element of the turn, and dJt is the element of

length along the coil. This equation becomes

<J> = nA / B • dl (2) c

under the following assumptions: (1) that the cross section A of the

coil is everywhere the same; (2) that the number of turns per unit

length n is constant; (3) that the B field is homogeneous over a scale

comparable with the dimension of a turn; (4) that the individual areas A

of each turn are oriented perpendicular to the curve described by the

coil. Hence, when c is a closed curve and when c itself links no flux,

the flux linked by the windings of the sensor is <J> = nAVIenc iose(j independently of c and the distribution of current density within it.

Thus, the enormous advantage of this geometry is due to its transportable calibration. An example of a Rogowski coil design appears in Fig- ure 1. The purpose of the housing is to guarantee that c encloses no

flux.

Rogowski coils in general have air or plastic cores such that

\i = the permeability of free space. The permeability of core materials drops with increasing frequencies [3], and for high frequency

applications an air core is adequate.

176
UNIFORMLY

CONNECTOR

Figure 1. A typical Rogowski coil. Aluminum housing shortcircuits major turn. Current viewing resistor is

built into output connector. Housing is cut to

allow flux to penetrate. Turns are uniformly spaced

2. 1 Lumped Parameter Model

A lumped parameter approach [7,8]

types of sensors when the transit time

to the resolution time of the metering

can be used to describe both

in the sensors is small compared

circuit. Figure 2 shows the

coil

-*/v»vR " RCVR + RCOIL

R CVR

> To

oscillograph

Figure 2. Equivalent circuit for field-coupled current

sensor. Sensor differentiates when t

-1 c integrates when t << LR

177

>> LR-1 and
circuit diagram of a field coupled sensor. Assuming no relative motion

between the current density distribution and the sensor, <f> is the time

rate of change of flux, linked by the sensor, due to changes of current

in the primary circuit. R = R

Q + R^-,^ is the sum of coil resistance R

and metering resistor R(~.y R

« L is the self-inductance of the sensor.

For this circuit,

154-Jifk +1 (3 ) R dt R dt c

where I is the current in the sensor.

c The three terms in the equation suggest two time domains. In the

first one the characteristic time t

c for changes in the primary current

is t

c >> L/R. In this domain, L/R dl

c/dt << Ic and the current in the

sensor is I - 1/R dcf/dt. In this domain, the sensor is differentiating

one since its output U = I C

R is equal to the <f> arising from current

changes in the primary circuit. The other time domain is t << L/R. In

this domain, L/R dl /dt > I . The current in the sensor is I - i> L~^ =

_ I

' C C _ 1 C

T IpN and the output voltage is U =

RcVR'^p1*' when Ip is the primary

current and N is the number of turns. This self- integrating sensor is

preferable to the differentiating one for reasons which will become

clear when we consider transit time effects in the coil itself. The

design of the self-integrating Rogowski coil is discussed in detail in

[7]. (The use of ferromagnetic cores in Rogowski geometry coils is

discussed by Anderson [3].)

2.2 Transmission Line Model

The lumped parameter circuit description of the field coupled cur- rent sensor is no longer adequate when the transit time in the sensor is

longer than the resolution time of the metering circuit. Large, fast

risetime pulsed power systems and the improved time resolution of metering circuits often require transmission line treatment of the field

coupled sensor (see Figure 3). The voltages induced at different locations along the sensor reach the metering impedance with different

delays. Transmission line treatment assures that transit time effects

are properly accounted for. In a Rogowski geometry, the two ends of a coil are joined so the coil forms a torus.

In 1963 J. Cooper [9] developed a transmission line model, shown in

Figure 4, for field coupled current sensors. His model includes the

distributed capacitance per unit length between the coil and the electrostatic shield, the distributed inductance per unit length , and the

distributed excitation voltage u(x) per unit length along the coil. He

applies a Laplace transform technique to solve the partial differential

equations that model the coil. The boundary conditions, which may be

applied to the sensor, include short circuit, resistive, inductive, and

capacitive terminations at the ends of the coil

.

178
c£dx

1)1/2

dr = (^C) 1/2 dx|

COCN

CN

Cdx

x+dx

Figure 3. Differential section of transmission line geometry

current sensor (u(x) is excitation voltage per unit

length)

CO00

CN

^SHIELD TRANSMISSION LINE IMPEDANCE = Z0 TRANSIT TIME = r \

Oscillograph

sCORE (m =Mq)

Cross-section at end COIL SHORTED

TO SHIELD

Plan view CURRENT

VIEWING

RESISTOR Z

Figure 4. Transmission line geometry current sensor. Gap in

shield allows flux to penetrate coil. Wave

impedance of coil is Zq

and its one-way transit time

is t. Coil is terminated with current viewing

resistor Z at one end and shorted at other end.

Coil forms closed curve for Rogowski geometry

We assume a delta function excitation to illustrate the effect of

transit time upon the sensor output. We assume I = 5(t-0)I such that

the primary current excitation is I = 0, t < 0 ; I = I, t > 0. The

excitation of the sensor is uniform, such that u( x; is a constant in the

Rogowski geometry sensor. The transmission- line- like sensor of impedance ZQ

, one-way transit time t, length I, with total number of turns N,

is terminated by a short circuit at one end and with a metering impedance Z at the other end. We measure the coil output as a voltage

across Z. The 6 function input and output for terminations Z >> Zq

, Z = Zq

and Z << ZQ

, appear in Figure 5. Cases (a) and ( b) are the

transmission line model of the differentiating sensor. Indeed, for a differentiating coil /uro ^jdt = <j> = yQ

InA. Assuming no flux leakage

between the coil and the primary circuit L = pQAnN so that <f> = ILN~^.

179
We integrate the waveforms (a) and (b) of Figure 5 and using

UdLcy/ = T and (^C 1) 1 ^2 = Z we obtain the waveforms (a) and ( b) of

Figure 6. The mean value of jucoildt = ILN~ 1 for Z >> ZQ and the final

value ILN~ 1 for Z = ZQ for t >> t agree with the lumped parameter

description. The sensor has, however, introduced a time structure that

did not exist in the current waveform under measurement. (The output of

a differentiating transmission line sensor is illustrated in Figure 5 of

our contribution to Diagnostics for Fusion Experiments [10].

z

hi

crceU

dl_ dt II IS ( T

-

2t At

TIME r. s

6r St

H

.

_JO

>

D

h

D

oH

D

O

cco

2

UJ

00

I Z,

Ho

2 N

(a) Z»Zr

(b) Z = Z,

(c) Z«Z0 Figure 5. Response of transmission line coil to step input.

Waveforms (a), (b) and (c) correspond to a current

viewing resistor impedance Z >> Zq/ z = ZQ

, and

Z << Z respectively

130
2 IL N

l_L

N

(a) Z » Z,

I LN

CM Figure 6.

2r 8r

(b) Z = Z,

Integral of transmission line sensor waveforms.

Waveforms (a) and (b) correspond to (a) and (b) of

Figure 5 (Z >> ZQ and Z = ZQ

) . Response for t >> t agrees with lumped parameter description (L =

t = >£ee)

The advantages of the self-integrating sensor are clear when we observe waveform ( c) of Figure 5. This waveform exactly reproduces the

primary current, and the frequency components of its output are no higher than the components of the primary current. Indeed, it is

possible to construct a subnanosecond risetime sensor whose physical

size results in t ) 10 ns transit time. Another advantage of this

sensor is that its output is insensitive to electron input or photon

induced currents. In constructing such a sensor, it is critical that

the turns are wound uniformly, that the major turn is shorted with the

electrostatic shield, and a sturdy, nonreactive current viewing resistor

built. Solutions to these and other technical details appear in the

literature [6,11]. Sample waveforms from a large, self- integrating

Rogowski coil, which demonstrate fast risetime, are illustrated by

Pellinen [7].

2.3 Sources of Error in Field Coupled Current Sensors

The low frequency cut-off of a self-integrating current sensor is

produced by penetration of magnetic flux into the "core" of the coil.

This penetration is characterized by a time constant = LR~^. While a high permeability core will increase this time constant, the risk of

saturation is always present in a coil with such a core. Therefore, an unknown current should not be measured with a ferromagnetic core coil

whose ampere-second rating is unknown.

The time varying resistance of the winding due to skin effects and

to heating the current viewing resistor may introduce some error in the

determination of t^. This quantity is required for a numerical

compensation of the decay obtained with Equation (9).

181
Nonuniform excitation (u = u(x,t)) is the Achilles' heel of the

self-integrating Rogowski coil. It causes severe oscillations in the

output whose period is the two-way transit time in the coil and which

damp very slowly due to the high Q of the sensor. When a measurement of

total current is required, a terminated differentiating sensor (Fig- ures 5b and 6b) is preferable even though the risetime information is

lost. This information can always be retrieved from a smaller selfintegrating fluxmeter. In practice, we measure current in large devices

(r > 0.2 m) with three or four sensors positioned symmetrically in the

same axial location. This also measures the degree of symmetry in the

current distribution. Damping resistors can also reduce oscillation

[3].

Another limitation and source of error is the temperature rise due

to power dissipated in RcVR. The risetime in this resistor is

t = 0.25 pp~ 1 6 = 0.25 t , where p is the resistivity and 6 is the

thickness of the CVR material. The temperature rise AT of the CVR is

C

p

DiT = 0.25 ppf- [ij (4)

D is the density of the CVR material, and w is its width.

We conclude this discussion of field coupled sensors with a brief

overview of differentiating sensors. In general, they are easy to build

and provide a high output voltage per given area. Their disadvantage is

the large bandwidth their data acquisition requires (short cable runs),

and their susceptibility to arcing due to fast transients. Trivelpiece

[12] describes a numerical technique for compensating cable aberrations,

while McDaniel discusses fast- transient limitations [13]. The high

frequency response of these sensors is governed by LR~ , which must be

smaller than the timescale characterizing the current being measured.

Since the pulse begins at t = 0, there is always an interval in which

t < LR~^. During this interval, a differentiating monitor delivers the

integral of the waveform. The task of the designer, therefore, is to

minimize the interval. Random errors in measurement with these devices

are caused by stray charges, photon-induced currents, and capacitive

coupling.

The voltage drop developed at the junction between two flanges is a source of systematic error when differentiating sensors are mounted at

such a location. One example of this application appears in Figure 7

[14,15]. The voltage drop is added to and integrated with the signal of

the sensor. If the sensor is a groove of area, A at a radius, a and the

contact resistance between the flanges is R, then the error, A is

A = T5

^

. contact o

182
E

Figure 7. Flange mounted differentiating fluxmeter

For the sensor shown in the figure ( a = 1 0~ m, t = 10 s, A = 0.6 x 10~4 m

2

), R < 10" 5

fi for A < 0.05.

The finite resistivity of the loop material is another source of

error in this arrangement. This source is present even when the loop is

simply a groove. Error is introduced by a voltage drop along the loop

due to the primary current, and a change in loop area caused by the

penetration of the magnetic field in the loop material.

The error due to the voltage drop in the loop material is

A = — * =4 (6)

IR, b loop

where 6 = 2 Vt p~ ^ p/ and b is the side dimension of a square groove. We

obtain the above result by assuming that current penetrates a depth 6 at

time t. Lastly, the error due to the increase in area of the loop is

4_6

b

183

(7)
These three additive errors, which become more pronounced for longer

pulses, are larger for more resistive materials. The last two errors

are considered in more detail by Chu [16], who also presents a technique

for their numerical correction.

3. Current Viewing Resistors (CVRs)

The widespread use of flux-coupled current sensors has been justified by their ruggedness, reliability, accuracy, and excellent frequency

response. Recently, however, the application of these devices to magnetically insulated vacuum transmission lines (MITL) has become somewhat

limited. The large electrical stresses associated with high power

densities and fast risetimes, and the presence of space charge clouds in

the gaps, break down either the sensor or its shielding groove. Moreover, the groove introduces a substantial inductance penalty in accelerators with a large rate of current rise. These difficulties have led to

the development of alternate current measuring techniques, such as the

CVRs described in this section or piezoelectric transducers developed at

other laboratories [17].

3. 1 Modeling a CVR

The frequency response of a CVR is limited by: (1) diffusion of

the magnetic field in the CVR material; (2) the insertion inductance of

the CVR in the circuit; and (3) the transit time of electromagnetic

waves in the CVR itself. Magnetic diffusion is the most important

limitation; thin CVRs of small dimensions have little insertion inductance. A small dimension also circumvents the problem of electromagnetic

wave transit time.

The time response of a CVR has been treated in detail in the

literature [4]. Consider a tubular CVR with internal radius r, external

radius a, length I, CVR material thickness r-a = 6, resistivity p, permeability \i, and ratio of radii e = ra" 1

. We introduce a characteristic time t

c = )jp~ 1 6 2

, characteristic voltage U

c = IpP£(2iTa6)

-

^ where

I is the peak current in the circuit. We further assume a CVR with a thickness a << a. Then the output voltage U of such a CVR to a step

function input of magnitude I is approximately

1 -1/2 ~ 4t * U* = 2 ( Tit* ) e (8)

where t* = tt

c

~ 1 = tp(y6z

) -1 and U* = UU

c

~ 1 = 2UTra6(I

p

p£) • Equation 8

describes a CVR response that is indistinguishable from the exact solution in the interval 0 < U* < 0.95. Equation 8 appears as the solid

line in Figure 8.

184
The diffusion process may also be modeled with an equivalent

network of inductors and resistors [18]. The circuit diagram of such a 10-section network appears in Figure 9. The inductance of each section

L is normalized with respect to the total inductance of the foil: L =

n _ -I _ I \ilaw such that L* = 0.1=LL where L is the thickness of each

n n n section and w = 2i\a. The resistance of each section is normalized with

respect to the total resistance of the foil: R = pJt(w6)

-1 such that

R* = 10 = LL

~ 1

. The time is normalized with respect to t = LR

-

^ =

2 -1 n c yo p As before, the output is normalized with respect to U .

Figure 9. Equivalent network model for current diffusion. The

input current is I* = I I ~ 1

; the output voltage

is U*

P

185
The result of a simulation using a current waveform with a risetime

-2 -2 of 5. 2 x 1 0 t ( t* = 5.2 x 10 ) appears as the dotted waveform in

Figure 8. This simulation waveform is indistinguishable from the ana- lytic approximation represented by Equation (1). Both the analytic approximation and the circuit analysis simulation show a 10 to 90 percent

risetime t* = 0.25.

3.2 Sources of Measurement Error

There are several techniques to compensate for the diffusion of

current in the CVR. One technique [4, 19] relies on a loop that links

some of the flux within the resistive material. Another taps the output

midway in the resistive material [20] by using a shunt made with two

parallel foils. A third technique relies on compensation with a passive

network [18,21].

To reduce error, the CVR should be a four- terminal network so that

the metering contacts are independent of the current-carrying contacts.

We also require a sturdy support for the foil. Figure 10 schematically

illustrates a CVR construction that accomplishes both objectives. The

foil (a) is stretched over a metal yoke (b). The metering contacts are connected to the foil at ( c)

.

TO OSCILLOGRAPH

CO

Figure 10. Cross—section of CVR. Foil (a) is stretched across metal yoke (b); metering contacts are at (c).

Current flows in direction of arrows

186
The yoke carries the current for pulse durations, which are long

compared to t^ = LyR

-

^ , where L is the inductance of the region between

the foil and the yoke, which is crosshatched in Figure 10. A decay time

of t^ = 10~^ s is easily obtained since foil resistances are typically

1 ft, and inductances of 10~^ K result from reasonable geometries.

The decay associated with t^ is compensated with the algorithm

t

U = U + / U dt (9)

m

o where Um

is the compensated value of the measurement

.

There are three other possible sources of error when measuring cur- rents with CVRs . The first source is spurious signals that arise from

the direct coupling of the measuring instrument ( oscilloscope or digitizer) and the circuit carrying the current. We reduce the error by

introducing either a decoupling transformer or a large inductance

between the CVR and the oscillograph.

The second source of error is the time-varying flux coupled in the

connection to the CVR. The coupling arises from the CVR construction

illustrated in Figure 10. It is easy to show that the ratio of resistive voltage, U_ to inductive voltage <t> is:

c m

U A

c _ y

: A

d>r m

m

(10)

where A^ is the area between the yoke and the foil, and ^ is the area subtended by the metering connections. Therefore, we must construct the

CVR, minimizing the area subtended by the connections to the foil. A

Park shunt minimizes this error [22].

Joule heating of the foil is the third source of error. This

affects the CVR response by changing foil resistance during the pulse.

Heating takes place in two phases. In the first phase, heating occurs

in the foil as the magnetic field diffuses in the conductor. In the

second phase, the current has fully diffused and heating takes place

throughout the foil.

We now discuss foil heating during the diffusion phase. Assume a step current I impressed on the foil at time t = 0. The penetration of

P —11/9 the current in the foil at time t is 5(t) = (2pty ) ' . The factor 2 arises from full penetration at tt ~ = t*/2. The resistance of the

foil is R(t) = p&(w6) • The energy input at time t is

t

2 f 2 pi nr~ 1/2 , ... E(t) = I / R(t) dt = 21 -J- J t . (11)

p I o w y 2p

187
This heat input occurs in a volume V = 6£w. Equating the thermal energy

of the foil with the energy dissipated in it, we obtain

C DAT = y I 2 w~

2 = 2B2 poo /2u (12)

o

where AT is the temperature rise of the foil, B = \iQIw~^ , and D is the

density of the material. This result is well known and originated from

work with pulsed magnetic fields [23]. It was found during the diffusion phase that the thermal energy per unit volume of the material is

equal to the energy density of the magnetic field.

Once current diffusion is complete, the thickness of the foil

carrying the current is constant. A parallel analysis to the one above

yields

C DAt = — ^ . (13)

P t 2y

c o

Both equations have a similar form. Making the foil thicker does not

affect the temperature rise in the diffusion phase. However, when

diffusion is complete, the thicker foil will have a smaller temperature

rise. Unfortunately, this is achieved at the cost of CVR response.

3.3 A Practical CVR Design

The CVR for the Macrotor REB drive experiment [24] appears in

Figure 11. The schematic of this CVR appears in Figure 10. Its dimensions are: a = 1.34 x 1 0~ 2 m; % = 7. 6 x 10~ 3 m; 6 = 7.6 x 10~ 5 m. Therefore, t

Q = 9.6 x 10~9

s, Uc /I = 9.01 x 10~ 4 V A~ 1 and the risetime

(t/t

c

= 0.25) is 2.5 ns.

CO

rCN

CO

Figure 11. CVR for Macrotor MITL

133
We measured the response of this CVR to a 500 ps risetime current

drive of 20 A with a sampling scope at a 50 0 ps/div sweep speed. The

response normalized with respect to t and U

c appears as the solid line

in Figure 12. If we compare the response to the simulation results in

Figure 8, there is significant oscillation (~ 1 GHz) in the leading edg

of the CVR output, not visible on a 250 MHz bandwidth oscilloscope.

Otherwise, the CVR output risetime of 2.5 ns compares quite favorably

with the numerical simulation.

K0.48 ns-H

P

D

Q.I-D

O

cc>

o

CVR response

ation

fi

•

• •

' / \

•) • / • /

/ q" i t 1

18 mV

0.1 0.2 0.3

TIME, t*

0.4 0.5

Figure 12. CVR output as a function of time for a step current

input. Voltage U* and time t* are nondimensional

Figure 13 shows a measurement obtained with this CVR on a 0.95-mlong MITL. The solid line is the MITL input current, while the curve

annotated with diamonds is the MITL output current. The input current

was measured with a conventional Rogowski coil; the output current was

139
measured with the CVR described above. The measurements were obtained

with a short circuit at the output end of the MITL. The two waveforms

have been displaced by the one-way transit time of light in the 0.95-mlong structure. The current measured by the CVR faithfully reproduces

the input current measured by the Rogowski coil, except for a slight

expected erosion of the leading edge of the pulse. The waveforms are almost identical in the vicinity of peak current.

The maximum linear current density in this CVR has been as high as

200 kA/8.4 cm = 24 kA cm" 1

, which implies a peak magnetic field of

3.0 Wb-m~^ at the surface of the foil. A temperature rise of about 5 °C

accompanied by less than 0.5 percent change in resistance is implied for

this particular CVR at peak current.

100

150

TIME, ns

Figure 13. Current measurements in a 0.95-m-long MITL. Input

current was measured with a Rogowski coil. Output

current was measured with the CVR shown in Figure 3

4. Calibrating Rogowski Coils, Fluxmeters, and CVRs

The calibration of current sensors is very difficult in the

> 100 kA current range. The difficulty arises from lack of sources with

a precisely known current. In this section, we discuss the relative

merits of various calibration methods.

From the standpoint of high frequency response, a cable pulser with

a reed switch [25] is the best source. Its advantages are: exactly

known output level, < 500 ps risetime, and reproducible repetitive

190
output. Its main disadvantage is its low level output ( ~ 20 A into a short circuit), which limits its application to bench calibration of

high sensitivity sensors (> 100 V/MA) . In situ calibration with this

pulser is even more difficult due to its low level output.

Another current source is a high voltage cable pulser. A spark gap

(air or oil, self-breaking, triggered, or mechanically actuated) con- nects the charge line to the load under calibration. This kind of pulser can drive as much as 400 A per cable. Unless it operates in the re- laxation mode or with a triggered spark gap, it is nonrepetitive . The

switching losses, which are somewhat difficult to quantify, and the

switch inductance become more important as we increase the number of

cables.

A third source applicable in the microsecond timescale is the ringdown of a capacitor in an RLC circuit. The current is easily obtained

with some simple calculations and with measurements of voltage, period

and decay rate.

A useful device for calibration is a thin foil shunt [26] which is

well characterized in the laboratory under dc and pulsed conditions, and

is used for calibrations in situ . A coaxial device of this type has

been used standardly in our laboratory for several years.

191
5. Summary

The design of current sensors depends heavily on the environment

and time domain of operation. There is no current sensor ideal for

every application. This paper points out a rationale for proper choice

of current sensor.

I am grateful to Don Pellinen—friend, former colleague and mentor.

This paper reflects many of his ideas and designs. I also thank

S. Stillwachs for the generous hospitality at Carnelian Bay, California,

where I wrote this paper. Thanks also to Anne and little Emily for

their patience, and to Margaret Thomas and Bronya Feldmann for their

skillful editing.

[I] S. Glasstone and R. Lovberg, Controlled Thermonuclear Reactors , D. van Nostrand, Princeton, NJ (1960).

[2] W. Rogowski and W. Steinhaus, Arch. Elektrotech. J_, 141 (1912).

[3] J. M. Anderson, Rev. Sci. Instr., _42, 915-926 (1971 ).

[4] R. Malewski, Rev. Sci. Instr., 39, 90-4 (1968).

[5] A. Ebinger, Archiv fur technisches Messen (ATM), 320-F4, 467 , 211-214, DK 621.317.31 (1974).

[6] D. Bartholomew, Electrical Measurements and Instrumentation , Allyn

and Bacon, Boston (1963).

[7] D. G. Pellinen, et al., Rev. Sci. Instr. _51_, 1535-46 (1980).

[8] F. Labuhn and K. Weinhardt, DK 621.31 7.31.014.33 (1971).

[9] J. Cooper, Plasma Physics, 5_, 285-9 (1963).

[10] M. Di Capua and D. G. Pellinen, in Diagnostics for Fusion

Experiments , E. Sindoni and C. Wharton, eds., Pergamon Press,

Oxford (1979).

[II] D. G. Pellinen, Rev. Sci. Instr., 42, 667-70 (1971 ).

[12] C. Trivelpiece, paper read at Workshop on Measurement of Electrical

Quantities in Pulsed Power Systems, NBS, Boulder, CO., March 2-4,

1981 (unpublished).

[13] D. H. McDaniel, ibid.

6. Ac knowl edgement

s

7, References

192
[14] C. A. Ekdahl, Rev. Sci. Instr., _51_, pp. 1645-48 (1980 ).

[15] , ibid., 1649-51 (1980).

[16] E. Chu, paper read at Workshop on Measurement of Electrical

Quantities in Pulsed Power Systems, NBS, Boulder, Co., March 2-4,

1981 (unpublished).

[17] R. R. Williams et al., IEEE Conf. Record, 14th Pulsed Power

Modulator Symposium, 80CH1573-5, ED, NY (1980).

[18] A. J. Schwab, IEEE Trans, on Power App. and Syst., PAS-90 , 5, 225-7

(1971).

[19] R. Malewski, IEEE PES Summer Power Meeting, Minneapolis, MN,

July 13-18, 1980; also IEEE Trans, on Power App. and Syst., PAS , (to be published in May 1981).

[20] , ibid., PAS-96, 2, 579-85 (1977).

[21] G. Hortopan, V. Hortopan, Rev. Rum. Sci. Tech., Electrochn. et

Energie, 19, 1 , 41-47 (1974).

[22] J. Park, J. Res. Natl. Bu. Std. _39, 191 (1947).

[23] H. Knoepfel, Pulsed High Magnetic Fields, Anerican Elsevier

Publishing Co., Inc., New York (1970).

[24] V. Bailey et al . , Japan-U.S.A. Compact Toroid Workshop A9, Osaka,

Japan, February 17-19, 1981; also Physics of Fluids (to be

published)

.

[25] R. L. Garwin, Rev. Sci. Instr. 21 , 903 (1950).

[26] D. G. Pellinen, ibid. 4JU 1347-8 (1970).

Section 8. Sponsor Acknowledgements

This work was supported partially by DNA, U.S. DOE and Physics

International internal funds.

193
PRECISE MEASUREMENT OF CURRENT IN PULSED POWER SYSTEMS

Michael E. Wilmer and Paul A. Pearson

Pearson Electronics, Inc.

4007 Transport Street

Palo Alto, CA 94303

In order to precisely measure currents in a pulsed power system, the

capabilities and limitations of the instrumentation used must be well

characterized. In this presentation, we shall examine the theoretical

and empirical properties of internally terminated current monitors in

pulsed power applications. The advantages of current monitors over current viewing resistors will be discussed. The factors affecting the inherent accuracy of current monitors will be analyzed and their temperature dependence reported. The effect of the current monitor on the circuit being measured will be examined and the insertion inductance and insertion resistance will be calculated as a function of the key current

monitor design parameters. The voltage standoff characteristic of these

monitors will be derived and the results tabulated for several clearance

hole sizes both in air and in transformer oil. Core saturation will be

related to the design parameters of the individual current monitors and

its effect on the output waveform will be examined. The talk will con- clude with an explanation of the usefulness of dc biasing in extending

the range of current monitors, particularly those employing high permeability cores.

Key Words: Current measurement; current monitor; current transformer.

1. Introduction

The accurate measurement of currents in pulsed power systems is

often crucial for the successful operation of the system. In applying

current transformers to this task, it is important that the limitations

imposed by the design constraints of the transformer be observed. In

this paper we shall review the theory of operation of terminated current

monitors and relate their performance characteristics to the key design

parameters. In particular we shall carefully investigate the various

factors which can lead to discrepancy between the actual primary current

waveshape and the secondary output voltage waveshape.

194
2. Competing Approaches

Current viewing resistors or resistive shunts are presently the

most practical alternative to the use of current monitors in measuring

pulsed currents. While the resistive technique is frequently adequate,

current monitors have several distinctive advantages which offer significant improvements in accuracy and' overall performance.

Since current monitors are transformers, the primary and secondary

are electrically isolated and interference and noise problems associated

with ground loops can be avoided. A further advantage of this isolation

is in the improved protection of personnel and equipment which it provides. There are two further disadvantages in the use of resistive

shunts for measuring pulse currents. First, resistive shunts are subject to an increase in resistance with frequency due to skin effect re- sistance [1,4] causing significant error in the measurement of pulses

with short rise and fall times. Second, resistive shunts present a relatively high resistance to the primary circuit. For a particular sensitivity S, a resistive shunt presents a load of S ohms to the primary

circuit while a current monitor presents a load of only S/N2 ohms where

N2 is the number of secondary turns. When the currents to be measured

are in the thousands of amps, this can be a very significant factor.

In the simplest case, a terminated current monitor consists of a magnetic core (typically toroidal in shape), a number of primary turns

Nj, a number of secondary turns N2, and a terminating resistor Rj.

Figure 1. Simplified Current Monitor

Normally, there is but a single primary turn and therefore the output voltage for an alternating primary current is:

3. Simplified Theory of Operation

V T = i 2 RT = U

"RI

N2

[1]

195
The ratio of Rj to N2 is called the sensitivity of the monitor and

is expressed in units of volts/amp. The equivalent circuit consists of

a pulse current source shunted by the secondary inductance, shunted by

the winding resistance in series with the terminating resistance. The

voltage which is generated across the terminating resistance in response

to a stepfunction current in the primary is given in equation (2).

y T - ilEl e

' t/x

VT N2

where t = l_2/(Rw + Rt)

(2)

The decay of this exponential term causes the output to droop. The

initial rate of droop (in percent per microsecond) is a parameter which

is commonly used in specifying current monitors.

The total voltage per turn is given by the sum of the voltage

dropped across the winding resistance (Vw) and the voltage dropped across

the terminating resistance (Vj). The magnetic induction B of the core is

equal to the integral of voltage per turn per unit area as shown in

equation (3).

B(t) = (Vt + Vw ) dT

o J N2 A (3)

For a step function excitation, we can substitute equation (2) into

equation (3), yielding equation (4).

= Ii(RT + Rw ) T(l-e" t/T

) A U2 Z (4)

For times which are short relative to t, the exponential may be

approximated by the first term of its power series expansion and B becomes

:

B = Ii( Rt + RW ) t (5)

A N2^

This equation shows that B increases linearly with time for times

short relative to the droop time constant, and after some time, B will

increase to the point at which the core is effectively saturated. The

product of primary current and time for which this saturation occurs is

another common parameter of current monitors and is called the currenttime product.

Iltmax = Bmax A N2 2

(f>]

(RT + Rw)

(6)

196
4. Factors Affecting Accuracy

When a current monitor is used to measure pulsed currents, there

are several factors which contribute to the potential error. If an ideal square pulse with an amplitude I 0

is assumed as the primary current, then the actual output of a current monitor with sensitivity S

would have the general shape shown in Figure 2.

IDEAL

SECONDARY

VOLTAGE

Figure 2. Ideal and Actual Waveform

There are four ways in which the actual output differs from an idealized output. These are (1), the rise time and fall time are not

zero; (2), there is some overshoot and ringing at the leading edge;

(3), there is some droop of the flat top with time; and (4), the output

voltage may not equal the ideal value of SI 0 because of sensitivity

error in the current monitor. We shall examine these errors one at a time in order to estimate their potential impact and to understand how

to minimize or compensate for each.

4.1 Rise Time Limitations

When an ideal step function of current amplitude \\ is passed

through the center of a toroidal current transformer having secondary

turns, a current 1 1/N2 is induced in the secondary winding. Several

authors have developed the high frequency equivalent circuit for terminated current monitiors [2,3] and there is general agreement that the

leakage inductance is negligible and that the rise time of the output

voltage waveform is limited by the distributed capacitance in parallel

with the terminating resistance. There is also a small inductance asso- ciated with the termination network which can give rise to ringing. Be- cause of the excellent high frequency response of terminated current

monitors, increases in observed rise time when the monitor is placed in

the circuit are usually found to be due to the inductance of the primary

connection.

197

ACTUAL

SECONDARY

VOLTAGE
Figure 3. Current Monitor input (lower

trace) and output waveforms with an 8

inch primary wire connection. Time

scale lOns/div.

In Figure 3, we see the waveshape which results when a 0.1 amp current pulse with a rise time of 5 nanoseconds is passed through a Pearson

Model 4100 current monitor using approximately 8 inches of wire. The

rise time of the resulting output waveform is approximately 7.5 nanoseconds.

Figure 4. Coaxial Current Monitor

test apparatus with a Pearson Model

4100 Current Monitor.

198
In an effort to minimize the effects of primary circuit inductance,

the coaxial current monitoring cavity shown in Figure 4 was constructed

to hold the model 4100. The response of the Model 4100, enclosed in this

device, to the same excitation which was used in Figure 3, is shown in

Figure 5. The rise time is approximately five nanoseconds. Thus, the

inherent rise time of this system is limited by the geometry of the primary circuit.

Figure 5. Current Monitor input (lower trace) and output

waveforms using the coaxial structure. Time scale 10 ns/div.

4.2 Overshoot and Ringing

Referring once again to Figure 5, we observe an overshoot of about

30% and several cycles of damped oscillation having a period of about 8

nanoseconds.

This phenomenon is due to a very high frequency complex pole in the

secondary circuit which arises from the distributed capacitance of the

secondary winding and the inductance of the resistive termination. Be- cause this pole is of such a high frequency in this particular current

transformer, its resonance is excited only when the rise time of the

primary current is short relative to the resonance period. It is generally true that large capacity current transformers have more secondary

turns and larger overall size and therefore have more distributed capacitance. For this reason, the resonance tends to occur at lower frequencies for larger units with lower sensitivity.

The usable rise time specification which is given for all standard

Pearson current monitors is that rise time for which the secondary reso- nance will be only negligibly excited when the primary current is passed

through the center of the transformer. It is also true that the rise

time of the primary current can be accurately measured if it is greater

than the quoted usable rise time.

199
4.3 Droop

As explained earlier, the droop of a current transformer is a direct result of the fact that it is a high pass device and when the droop

rate is known, pulse measurements can be corrected for droop. This is

most easily done when the total amount of the droop is small (less than

10%) so that the exponential decay can be accurately estimated by a linear decay. Thus, for a transformer with a droop rate of a%/ysecond,

the corrected output voltage at a time T following the onset of the

pulse is given in equation (7).

Vc(T) = Vqut(T) (WStH (7)

The correction given in equation (7) is most accurate for times

short relative to the droop time constant and for ideal square pulsed

currents, but it also can be used to reduce error due to droop for pulses

having other shapes.

4.4 Sensitivity Error

Sensitivity error in current transformers is most readily observed

when the current pulse being measured has insignificant rise time, overshoot, or droop error. It has two causes which can be readily understood

upon examination of equation (1). There can be a difference between the

specified number of secondary turns and there can be a difference between

the specified value of the terminating resistance and its effective value.

The number of turns can be very carefully controlled in manufacture

and therefore only the resistance error is significant. This effective

resistance is actually composed of the physical terminating resistor network shunted by a resistance representing the hysterisis and eddy current

losses of the transformer core. These losses are typically small and

therefore their equivalent shunt resistance is fairly large. For most

practical devices, this resistance contributes an effect which is between

0% and 0.5% and therefore, even though it exhibits some frequency dependence, its impact on overall accuracy as a function of frequency is quite

smal 1

The conclusion which can be drawn is that the sensitivity error is

primarily attributable to the secondary termination resistance. The sen- sitivity of the device can be determined by comparing its output to the

voltage dropped across a wideband precision resistance standard driven

by a pulse current source. The resistances used in Pearson current

monitors are constructed of extremely stable Manganin wire with a very

low temperature coefficient of resistance. This results in current

monitors whose error is essentially independent of temperature over the

operating range.

200
5. Effects on Measured Circuit

The effects of a current monitor on the circuit it is monitoring

can be divided into indirect effects and direct effects. The indirect

effects are changes to the circuit layout necessary to make the connection through the aperture of the transformer. The most common and most

important phenomenon is an increase in inductance which leads to increased

rise time in the primary circuit. This effect can best be avoided by

minimizing the length of the conductors and otherwise choosing a geometry

which minimizes inductance.

The direct effect of a current monitor can readily be calculated

upon consideration of the equivalent circuit in Figure 1. The impedance

added to a single turn primary is essentially all due to the terminating

resistance reflected into the primary. The relationship governing this

transformation is given in equation (8).

RlN (RJ+RW) (8)

For reasonably designed current monitors, this value lies between

10"° and 2 x 10" 2 ohms. The winding capacitance and inductance of the

terminating network are extremely low and therefore have negligible effect on the primary circuit within the operational bandwidth of the cur- rent monitor.

6. Voltage Isolation

A current monitor enclosing a primary conductor forms a coaxial

configuration. In order to estimate the voltage at which breakdown oc- curs between the primary and the current transformer shield, we will calculate the breakdown voltage for an ideal, endless coaxial line.

The electric field within a coaxial line is given by equation (9).

The voltage is the integral of the electric field between the conductors

and is given by (10). In this equation, a is the radius of the inner

conductor, b is the radius of the outer conductor and e is the dielectric

constant of the medium.

E = p/2ttc (9)

V = pin (!•) 2tt£ (10)

a

The maximum value of E occurs at r = a and therefore the voltage as

a function of inner conductor radius, outer conductor radius, and maximum

field strength is given by (11).

201
V = EMAX a In h (11)

a

Maximizing V as a function of a, one finds that the maximum voltage

occurs when In (b/a) = 1 or when a = b/2.73, and for this case, the maxi mum voltage is given in (12).

V = a EMAX (12)

In Table 1, we show optimum values of inner conductor diameter and

resulting calculated breakdown voltages in air and in oil for several

current monitor hole sizes. These assume breakdown strengths of 80 kV/

inch for air and 400 kV/inch for oil. Experimental work has shown these

numbers to be quite conservative.

Table 1. Voltage Breakdown

Current Approx. Optimum Approx. Breakdown Approx. Breakdown

Monitor Primary Conductor Voltage Voltage

Hole Diameter Diameter In Oil In Air

(Inches) (Inches) (kV) (kV)

2 .75 150 30

3.5 1.25 250 50

10.75 4 1000 200

7. Core Saturation Effects

Due to the limiting values of saturation magnetization in ferromagnetic materials, all transformers built around ferromagnetic cores are subject to saturation. Saturation can be defined in several ways, but

perhaps the most meaningful observation is that the effective permeabili

ty of the material rapidly decreases with the onset of saturation. This

in turn results in a dramatic increase in the droop rate for pulses.

For terminated current monitors, saturation can be predicted for

pulses short with respect to the droop time constant by computing the

current-time product. This product is actually the integral of the

primary current pulse with respect to time and for rectangular pulses,

it is simply the product of pulse width and amplitude.

In the situation in which a current monitor is being operated near saturation, the increased droop can cause error in measurements at or near the trailing edge of the output pulse. Since saturation is actuall

a voltage effect, reduction of the terminating resistance of the current

transformer effectively increases its current-time product. This can be

accomplished by externally terminating the current monitor with a 50 ohm

202
termination (note that this reduces the output to half of nominal value.

If you are using a device somewhere close to its saturation ra- tings, it can be quite convenient to quickly decide whether or not the

pulse current is causing saturation. One way to do this is by looking

at the waveform with the current monitor terminated in a high impedance

and then looking at it with the device terminated in 50 ohms. A significant difference in the latter part of the pulse waveshape with the

50 ohm termination versus the high impedance termination is an indication that the device is being operated at or near saturation.

The effective current-time product for a given device can be more

than doubled through the use of DC biasing. When a DC current of polarity opposite to the pulse is passed through the secondary of a current

transformer, the effect is to move the operating point of the magnetic

core to the opposite side of the hysterisis loop, thereby providing a much larger change in magnetization before saturation.

References

[1] F.D. Bennett, and J.W. Marvin, "Current Measurement and Transient

Skin Effects in Exploding Wire Circuits," The Review of Scientific

Instruments , Vol. 33, pp. 1218 - 1226 (19621

[2] F.J.U. Ritson and J. Wood, "Wide Band Current Transformers for the

Examination of Current Waveforms," Electronic Engineering , pp. 483 - 485 (July 1964)

[3] J. Cooper, "On the High Frequency Response of a Rogowski Coil,"

Plasma Physics (Journal of Energy, Part C ), Vol. 5, pp. 285 - 289

(1963)

[4] John H. Park, "Shunts and Inductors for Surge Current Measurements,

Journal of Research, National Bureau of Standards , Vol. 39, pp. 191

212 (1947)

203
LOW- INDUCTANCE SHUNTS FOR MEASURING LARGE PULSED CURRENTS

Walter F. Praeg

Accelerator Research Facilities Division

Argonne National Laboratory

Argonne, IL 60439

Precision shunts have been developed at Argonne National Laboratory

(ANL) to measure pulsed currents. Direct-coupled air-or water-cooled

coaxial shunts have been built for a frequency response up to 100 kHz

and with an rms current rating of up to 10 kA. Smaller shunts were designed to terminate current transformers for measuring <. 300 kA pulses

of 50 to 3000 ys duration. Some of our applications require the shunt

resistance to be repeatable within ± 0.005%. This made it necessary for

us to determine the resistance-stress coefficients of manganin and to

design the shunts for minimum stress. Stress in the manganin is caused

by thermal expansion, by cooling-water pressure, by mounting arrangements, and by magnetic fields. Design details and performance data are presented

.

Pulsed magnets used with the beam lines of the Zero Gradient Synchrotron at ANL have peak currents ranging from a few kA to several

hundred kA. The pulse width varies from 50 ys to 3000 ys. The required

stability and repeatability of current pulses was <. 1% for pion focusing horns, <. 0.1% for quadrupole magnets and <. 0.005% for dipole

switching magnets. These current pulses were measured with low inductance shunts which were either directly in the current path or coupled

to it via current transformers (CT's). No suitable shunts were available commercially, we, therefore, developed precision coaxial shunts.

The EMF between shunt potential leads may be written as

1. Introduction

E = I(R + jwL) + d<()/dt (1)

where

R

=

I = shunt current,

2irf

,

effective resistance of shunt,

204
L = four- terminal inductance of the potential circuit, linked with

flux set up by shunt current embraced by potential circuit,

d<f>/dt = voltages induced in potential circuit due to flux of currents

flowing in circuits other than the potential circuit.

The effective shunt resistance will change as a function of • frequency of shunt current, • temperature, • time, • chemical reactions with surroundings, and • stress.

For our applications, coaxial shunts were chosen because • the four- terminal impedance can be computed from the dimensions , • uniform stray fields will not affect the potential circuit, • there is a nearly uniform temperature distribution throughout

the resistance element, • they are simple to water-cool.

A coaxial shunt consists of a pair of concentric cylinders. One

cylinder, or part of it, is made from resistance material, the other is an external or internal copper cylinder. These cylinders are joined on one end by an annular conductor; the other ends connect to current

terminal s

2. Effective Impedance of Coaxial Shunt As A Function of Frequency

The impedance of a coaxial shunt has been computed by Silsbee [1].

His formula accounts for the current distribution over the thickness of

the coaxial conductors due to skin effects. It applies as long as the

potential leads are connected to the outside of the outer cylinder or to

the inside of the inner cylinder. Simplifying the exact solution to cover frequencies above 1 kHz and using the unprimed dimensions of

Figure 1, the impedance of the inner cylinder is

7 = d x (sin 6 cosh 6 + cos 5 sinh 6) + j(cos 6 sinh 6 - sin 6 cosh 6) /m

L Kdc6 (sin 6 cosh 6) 2 + (cos 6 sinh 6) 2 1 1

For low frequencies, the ac resistance of the shunt is [2]

(3)

and the inductance [3] is abhenries per cm is

205
L = - ^rd d

2 (4) 3b " 6b 7

where

6 = 2TTd/f7^,

d = b - a = thickness of shunt cylinder wall, in cm,

f = frequency of current through shunt,

Rdc

= ^ c res i stance of cylinder in abohm, and

p = resistivity of shunt material, in abohm-cm.

Equations (2), (3), and (4) also apply when the resistance material is in the outer cylinder. In this case, the primed dimensions of

Figure 1 must be used.

Manganin was chosen because of its stability with time and its low

temperature coefficient of resistance. General information on this alloy is given in Table I.

Manganin cylinders are usually not commercially available. A

nearly homogeneous cylinder can be made by bending sheet material into a circle and butt-welding the joint, using tungsten inert-gas or electronbeam welding techniques. After assembly and before the potential leads are attached, the manganin has to be artificially aged to avoid a slow

decrease in resistance with time. Commonly this is done by baking the

shunt at 150°C for 60 hours. Within two or three months after baking,

the shunt resistance will ordinarily increase by about 0.01% and will

then be stable and ready for calibration [4].

Change of resistance versus temperature of commercial shunt manganin is illustrated in Figure 2. The temperature at which the maximum

resistance occurs is a function of the thermal and mechanical treatment

of the manganin as well as of its composition.

Figure 1. Horizontal cross section

of coaxial shunt

3. Choice of Design and Resistance Material

3.1 General

206
Table I. Physical Characteristics of Manganin

Nominal analysis 84 Cu 12 Mn 4 Ni

Coefficient of linear expansions

between -20O-100°C 0.000 018 7 °C _1

Resistivity at 20°C wire 48.2 yft cm shunt strip 38.1 yft cm Temperature coefficient of resistance

between 15°-35°C for wire + 0.000 008 °C_1

between 37°-57°C for shunt strip + .000 004 °C _1

Thermal EMF against copper between 0°-100°C < 0.003 mV °C_1

Approximate melting point 1020°C

Tensile strength at 20°C > 2800 kg cm

-2

Figure 2. Change of resistance versus temperature

of shunt manganin

Figure 3 is an assembly view of a coaxial shunt. The upper current

terminal, which is silver-brazed to the inner copper cylinder, supports

the weight of the lower current terminal and the upper extension cylinder through spring assemblies. An 0-ring seals the cylinders. The

shunt is cooled with distilled water, which is de-ionized, de-oxygenated,

and filtered. Corrosion of the manganin due to cooling water is re- tarded by an epoxy-phenol ic coat of about 0.005 cm thickness. A fiberglass jacket 0.8 cm thick, which is separated from the manganin tube by

a layer of silicon grease, protects .the manganin.

3.2 Potential Leads

The effects of uniform magnetic stray fields on the potential circuit are reduced by placing four pairs of potential leads symmetrically

around the manganin cylinder and connecting them in parallel. Since

these leads cannot be attached on an equi potential line, a circulating

current will flow through them. This current is kept small by making

these leads from flat-rolled manganin wire which gives each lead a re- sistance of a few tenths of an ohm. The leads are insulated from and

taped to the manganin cylinder.

207
Figure 3. Assembly view of

coaxial shunt

3.3 End Effects

There are no end effects in a coaxial shunt if the current in the

cylinders is distributed uniformly around the axis and if the magnetic

field is not affected by the current in the current terminals.

These conditions might be realized by extending the current carrying cylinders, which connect to the current terminals, several diameters

beyond the location of the potential leads. In our case, this would

have resulted in very tall shunts. To keep the height down, we forced

uniform current distribution by cutting slots in the current terminals,

as shown in Figure 4.

The separation of the current terminals and their current distribution (eddy current effects) affects the voltage drop across a pair

of potential leads. These induced voltages have been computed [5];

208
they cause an error of < 0.001% in our application

Figure 4.

View of manganin

cylinder, current

terminal and lower

extension cyl i nder

,

Figure 5 gives |Z|/Rj~ and Rac /Rdc computed from Eq.(2) for a 200 \iQ

shunt rated 10 kA (shunt A), and for a 175 yft shunt rated 6 kA (shunt B).

Some features of these shunts are listed in Table II.

Table II. Data on Water-Cooled Coaxial Shunts

Shunt Type A

,

B

Current rating 10 000 6000 A

Resistance between potential leads 200 175

Length of manganin cylinder 30.48 30.48 cm Length of manganin cylinder between

potential leads 25.3 24.5 cm Diameter of manganin cylinder (ID) 24.15 14.8 cm Thickness of manganin cylinder 0.0635 0.1143 cm Heat flow in manganin cylinder 10.5 5.54 W cm

-2

Water flow 3.8 5.2 Liter s -1

Radial water pressure <0.36 <0.47 kg cm

-2

Temperature of manganin above water

temperature (20°) 40 17 OC

Overall heights of shunt 98 76 cm

3.4 Effects of Transverse Magnetic Fields

The electrical resistance of metals changes in magnetic fields.

For the transverse magnetic field caused by the shunt current, the in- crease of electrical resistance of manganin is given by [6], [7]

AR

R

= K

B

2 3B. (5)

209
where

aR = change in resistance,

R = resistance at zero magnetic field,

K = 1.16 x 10- 5 gauss" 1

, B = transverse magnetic field, in gauss, and

B

c

= 1.4 x 10 5 gauss.

The peak transverse magnetic field in the cylinder of shunts A and B at

10 kA are 166 G and 270 G respectively. These fields cause a negligible

resistance increase.

4. Changes in Resistance Due to Stress

Stress in manganin can change the shunt resistance beyond tolerable

limits. Unfortunately, manufacturers of manganin do not include resistance-stress coefficients in their general information nor could they

give such information upon request. A subsequent literature search re- vealed that several investigators had measured resistance-stress coefficients of thin manganin wires. In these measurements, stress was applied in the direction of current flow, which is not always the case with shunts.

Below the proportional limit, which we measured as 1050 kg cm

-2 on anneals strip samples, the resistance changes linearly with stress.

210
The dependence of resistance upon stress is

R

a

= RQ

(1 + aa) (6)

where

= resistance under stress and ambient temperature,

R0

= resistance under zero stress and ambient temperature,

a = (1/R) (dR/da) = resistance-stress coefficient,

a = eE = stress,

e = strain, and

E = Young's modulus of elasticity.

4.1 Resistance Changes Due to Stress in the Direction of Current Flow

From the literature [8], [9], and [10] a resistance-stress coefficient for tension in the direction of current flow was obtained as aji = 0.59 x 10~ 6 cm 2 kg" 1

. For pressure in the direction of current flow

the coefficient is aj^ = 2.31 x 10" 6 cm 2 kg -1

.

From Eq. (6) we obtain the relative change in resistance AR/R

Q

= aa. With a£ = 0.59 x 10" 6 cm2 kg" 1 and a" = 2.31 x 10 -6 cm2 kg" 1

, the stress

that causes a 0.01% change in resistance is

for tension a

t

= 10

_i+ /(0. 59 x 10" 6 cm 2 kg -1

) = 170 kg cm" 2

,

for compression, a

p

= 10~V(2.31 x 10" 6 cm 2 kg" 1

) = 43.3 kg cm

-2

.

4.2 Resistance Changes Due to Stress Transverse to the Current Flow

4.2.2 Tension

Radial pressure is exerted on the manganin cylinder of Figure 3

by the cooling water causing tension in a direction transverse to the

current flow. Resistance coefficients for transverse stress could not

be found in the literature and were, therefore, measured on the shunt of

Figure 3 with the circuit of Figure 6. In this circuit, after standardization of potentiometer P-^ , the voltage drops across the reference re- sistor R

s and the coaxial shunt R

x were compensated to zero. The value

of R

x

is then computed from the potentiometer readings and the known

value of R

s

. Corrections for variations in current between the successive measurements of R

s and R

x with Pi were made by monitoring these

fluctuations with potentiometer P

2

- Measurement accuracy was 0.02%.

The relation of resistance to radial pressure in the manganin cylinder of the 200 yft coaxial shunt was measured three times, and the

resistance found to be a linear function of stress within an experimental error of ± 4%. From this the resistance-stress coefficient for

tension, transverse to the current flow, was calculated as

211
a-j: = -0.78 x 10 -6 cm 2 kg -1

. The water pressures for shunts A and B are

<_ 0.36 and <. 0.47 kg cm

-2

, causing a change in resistance of >. -0.0053

and >. -0.0023%, respectively.

LING POWER SUPPLY

O.O! %

Sll

ST2' wr

W2'

Kin-Tel Electronic Galvanometer, Model 204A

Kin-Tel Electronic Galvanometer, Model 204A

L & N Universal Potentiometer K3

L & N Universal Potentiometer K3

Water Cooled Resistor

6KA Coaxial Shunt No. 3 (174.92/ifl)

Coaxial Shunt to be measured.

Standard Cell Type 1268, 1.01859V abs. at 20°c

Standard Cell 1.01927V abs.

Working Battery 2V 200 Ah, Willard Type DD-5-1

Working battery 2V 200 AH, Willard Type DD-5-1

4.2.3 Compression

Figure 6. Resistance

measuring circuit

Circumferential pressure is exerted on the manganin cylinder of

Figure 7 by the coolant causing compression transverse to the current

flow. The resistance-stress coefficient for transverse compressive

stress was measured in the circuit of Figure 6 as a£ = 1.73 x 10~ 6 cm2 kg -1

.

WATER INLET

STRESS IN DIRECTION OF CURRENT FLOW

POTENTIAL LEADS

THERMOCOUPLE

WATER OUTLET

0" RING GASKET

. UPPER COPPER

EXTENSION

CYLINDER

COPPER CYLINDER

LOWER COPPER

EXTENSION

CYLINDER

COMPRESSION

STRESS TRANSVERSE TO CURRENT FLOW

/

1.6

1.4

I 2

10

8

64

.20

24

6

+

o

X ^00 400 600 800 / 1000 TENSION^

Figure 7. Water-cooled shunt Figure 8 Resistance-stress

sensitivity of manganin

212
The resistance stress coefficients of manganin in cm2 kg -1

x 10~ 6 are:

For tension in the direction of current alji = 0.59,

For tension transverse to the direction of current a^ = -0.78,

For compression in the direction of current a£ = 2.31,

For compression transverse to the direction of current a£ = 1.73,

Figure 8 illustrates the resistance-stress sensitivity of manganin.

5. Sources of Stress

5.1 Longitudinal Thermal Stress

The thermal stress a induced in a restrained bar by a temperature

change AT is

a = Ea(AT) (9)

where

a = coefficient of linear expansion = 18.7 x 10~ 6 °C -1 for manganin

E = Young's modulus of elasticity = 1.39 x 10 6 kg cm

-2

A resistance change of 0.01% due to compressive stress of ap = 43.3 kg

cm"2

, is caused if we increase the temperature of a rigidly mounted

strip of manganin by

T 43.3 kg cm" 2

, A70r A1 1.39 x 10 b kg cm" 2 x 18.7 x 10 b UC" 1 "

i ' t)/ u *

The resistance change due to thermally induced compressive stress

is 15 times larger than the resistance change due to the temperature

coefficient of resistance. It is apparent that the resistance material of a precision shunt must be free to dilate. The part of the manganin cylinder of Figure 3 that is embraced by potential leads can expand radially, and the 0-ring allows vertical movement.

One coaxial shunt of nominally 200 yfi (Type A) and four of nominally 175 yft (Type B) were fabricated at ANL. Without cooling water,

their resistances were measured at the NBS at 30°C and 40°C with direct

current. Their change in resistance averaged 32 ppm or 3.2 ppm °C -1

which can be fully explained by the temperature coefficient of re- sistance of manganin.

For comparison, two commercial shunts were also measured. Their

resistance element consists of a number of spaced, paral lei manganin

strips secured at each end in massive copper terminal blocks. Their

limit of error is stated as 0.04% at 3 kA. The change in resistance of

these shunts over a temperature range from 35°C (1 kA) to 49°C (2 kA)

213
averaged 702 ppm or 50 ppm °C _1

. This is 16 times larger than what could

be explained by the temperature coefficient of resistance. This additional change is due to thermally induced compressive stress in the

manganin strips.

5.2 Pressure of Coolant

Radial pressure is exerted on the manganin cylinder by the coolant

(forced air, cooling liquid, etc.). The water pressures for the shunts

of Table I cause a change in resistance of <. -0.0053% for shunt A and

< -0.0023% for shunt B.

5.3 Magnetic Forces

The magnetic field produced by the current flowing through the co- axial cylinders will act upon the current itself. These magnetic forces

will exert pressure on the outside of the inner cylinder, tending to

shrink it, and pressure on the inside of the outer cylinder, tending to

expand it. Expressed in kg cm" 2

, these pressures are [2]

P

in

- 0.2162 x 10-312

p out

. o.2i62 x io-3 t2

(10)

(11)

where

I = shunt current in amperes, and

a ,b,a-' ,b' = cm.

For the manganin cylinders of shunts Type A and B, the radial magnetic pressures at 10 kA are 0.001 kg cm" 2 and 0.0029 kg cm" 2

, re- spectively.

6. Alternate Coaxial Shunt Designs

6.1 Manganin Cylinder Cooled on Both Sides

Shunts are often used in conjunction with water-cooled power

supplies. Usually the water is at static pressures too high for one- sided cooling. To relieve the manganin from static pressure both sides

must be exposed to the coolant. Figure 9 illustrates a design with both

sides of the manganin equally cooled. In the design of Figure 10 a few

holes have been provided in the extension cylinder and in the disk soldered to the top of the manganin to relieve static pressure; here cooling is done essentially on the outside of the manganin. The allimportant freedom of vertical expansion for the manganin is assured by

a copper braid assembly that connects the second current terminal to

the manganin via an upper extension cylinder (Figure 9) or via a copper

disk (Figure 10)

.

214
.Figure 9. Shunt with manganin Figure 10. Shunt cooled on outside

cooled on both sides. but without static pressure

on manganin

6.2 Air Cooled Shunts

Convection cooling can dissipate approximately 0.04 W cm

-2 at a temperature rise of about 15°C above a 20°C ambient. Forced air cooling

will increase the dissipation by up to an order of magnitude. An inexpensive design is illustrated by Figure 11. End effects are reduced

by arranging cables, to connect the shunt to the external buswork,

symmetrically around the axis of the shunt as shown in Figures 7 and 12.

Figure 11. Assembly view of Figure 12 Figure 13. 60 A

air-cooled shunt External connections shunt

215
7. Measuring Very Large Currents Via Current Transformers

Single sinewave current pulses of <. 300 kA and 167 Hz or 3.5 kHz,

into pion focusing horns, were measured by connecting two current

transformers (CT's) in series and terminating them with a coaxial

shunt. The focusing horn is the one-turn primary threading through

the 92 cm ID of the toroidal core of a CT with a 1:25 turns ratio.

This CT connects to a 1:200 CT which is terminated by the 500 yfi, 60 A

shunt shown in Figure 13.

8. References

[1] F. B. Silsbee, "Notes on the Design of Four-Terminal Resistance

Standards for Alternating Current," NBS Research Paper RP 133,

Vol . 4, January 1930.

[2] J. H. Park, "Shunts and Inductors for Surge-Current Measurements,"

National Bureau of Standards, Washington, D.C., NBS Research Paper

RP 1823, Vol. 30, September 1947.

[3] "A Study of the Inductance of Four-Terminal Resistance Standards,"

Bulletin of the Bureau of Standards, Vol. 13, pp. 375-421,

June 1916.

[4] J. L. Thomas, "Precision Resistors and Their Measurements," NBS

Circular 470, October 1948.

[5] W. O'Neill, "Error Voltages Induced in Potential Circuit of

Coaxial Shunt by Spatial Disparties in Current Terminal,"

Argonne National Laboratory, Argonne, IL, Particle Accelerator

Internal Report WD0-33, July 18, 1966.

[6] P. Kapitza, "The Change of Electrical Conductivity in Strong

Magnetic Fields," Proc Roy. Soc. (London), Vol. 123, pp. 292-

372, March 1929.

[7] Smithsonian Physical Tables. Washington, D.C.: Smithsonian Inst.,

p. 463, 1959.

[8] P. W. Bridgman, "The Effect of Tension on the Electrical Resistance of Certain Abnormal Metals," Am Acad. Proc, Vol. 57,

pp. 41-66, April 1922.

[9] J. 0. Linde, A. von Heijne, and E. von Sabsay, "A Study of the

Effect of Tension on the Electrical Resistance of Alloys,"

Arkiv fur Fysik, Bd. 2, No. 10, pp. 81-87, 1950-51.

[10] I. Vigness, "Investigation of Stress-Strain Relations of Meta

Wires by Electrical Resistance Changes," J. Appl . Phys., Vol. 23,

pp. 43-47, January 1952.

Work supported-by the U. S. Department of Energy

216
fllEASUR I NG FAST PULSE CURRENT USING LOW

INDUCTANCE CURRENT UlEWING RESISTORS AND di/dt PROBES

C. A. Hluehlenweg

Engineering manager

T !! (fl Research Products/ Inc.

Albuquerque/ New IBexxco 87108

G . lib Duff BSEE

Los Alamos Scientific Laboratory

Los Alamos, New fllexico 87545

1. Introduction

The ever increasing use of thyratrons and stripline pulse forming

networks (PFN) in pulsed circuits has created a need for an accurate, low

noise, extremely low inductance, wide bandpass method of current measurement.

Recently designed Current Uiewing Resistors (CUR) meet and surpass most of

the requirements. Quality diagnostics are essential in the development of

high efficiency, low loss, pulsed accelerators and a variety of other pulsed

powered circuits.

Research and development in the area of current measurements for

specialized requirements has been in progress for at least two (2) years.

The purpose of these efforts were two-fold. to develop a low inductance

CUR that could easily be integrated into fast pulsed thyratron circuits

and develop a method of directly measuring current rate of change (di/dt)

at the high bandwidth.

This report contains the results of these efforts and a simplified

description of how the CUR and di/dt probes were designed and fabricated.

2. CUR Characteristics

Typically three (3) characteristics of CUR's are of particuler

interest to the user: (1) input inductance, (2) frequency response and

(3) pulse energy capacity. Good estimates of the input inductance of

coaxial CUR's can generally be obtained from a summation of series coaxial

cylinder of inductance by:

. / . . / \ n . n -7 , . outer dxameter , L(nH/m) = 2x10 xnat log — 7— (U inner dxameter

Similarly the inductance of a flat plat CUR (stripline) may be computed

from

:

217
i / u \ on . separation betmeen plates (xn) _ ^ L , . N „ ,_ „ . , L(nHJ = 2Pxx—c — r

. , > —» -xlength (in)x2.54 (2) width of plates(xn) v v '

and, typically, ranges from a few to less than one nanohenry.

CUR frequency response is primarily determined by skin effect current

distribution xn the resistance foil. This mechanism was described in

detail by J. H. Park in an analysis attributed to F. B. Silsbee.

The important results of this analysis presented graphically in the

Park article in an impedance versus a scaled frequency parameter functionally

dependant upon foil thickness and resistivity.

Pulse testing some years ago experimentally confirmed the theoretical

analysis and all subsequent CUR design has been based upon this theory.

CUR frequency response generally ranges between 2000fIlHz and a few

10's of KHz and is typically coupled to the CUR energy requirements.

This latter parameter, as would be expected, is primarily dependant upon

the resistance foil mass.

It follows then, that the three (3) prime factors in CUR design (1)

inductance, (2) bandpass, and (3) energy capacity are interdependent and,

therefore, often necessitating design compromise.

3. Thyatron CUR

Resent advances in hydrogen thyratron technology have made these devices

available for a variety of applications. With the increasing use of the

thyratron in R X D experiments, the need for accurate diagnostics becomes

a necessity.

Current Uiewing Resistors are the most economical solution to the

problem of accurate current measurements. CUR's are easily custom tailored

to the wide range of pulsed currents of thyratron circuits. The output

signal of the CUR is clean and relatively noise free over several orders

of magnitude. Due to the extremely low source impedance, noise due to

E IT) I is reduced and the CUR is conveniently terminated into 50 ohms.

Several generations of CUR configurations were evolved in the program

and two (2) basic designs were eventually developed.

The standard version has an inductance of 35 to 40nH and for circuits

where inductance is critical, a low inductance unit of 3 to 5nH was developed.

Both versions serve as the mounting flange for the respective thyratron.

The standard CUR mounting flange is shown in Figure 1 and the low inductance

version is shown in Figure 2. A closer look at both figures reveal

several important features.

Note the removable di/dt probes wrapped around the base of the CUR.

These probes are discussed in detail later in this report.

Proper cooling is also an important feature to thyratron life and two (2)

218
methods of cooling are provided on each CUR.

Thyratrons that are operated in oil can utilize the open system

of cooling. This method directs oil onto the base of the thyratron and

returns it to the environment through an exit port.

The inlet port (seen in Figure 1 ) is the tower hose brad and the

exit port is the large square hole. For thyratrons operated at high power

where more cooling is required, a closedsystem for chilled water is also

provided. The closed system should be used for thyratrons operated in air

to maintain the cathode at a safe operating temperature. The closed

system cooling ring is directly beneath the cathode mounting flange as seen in Figure 1.

Figure 1. Standard thyratron CUR mounting flange

Figure 2. Low inductance thyratron CUR mounting flange

219
The standard version CUR was tested in the circuit shown in Figure 3.

CHARGE

INDUCTOR CHARGEDIODE FILTER

I ryyip—i—I—^ i i HRPP

*

I2W SOLID STATE LASER

LDM-I TDM5

LASER

TRIGGERED

SOLID STATE

DRIVER

BIAS

SPECIAL <

THYRATRON >

CVR

]

LOAD

CVR

COMMON POINT GROUND

Figure 3. CUR test ciruit

Maximum test parameters for the CUR test circuit in Figure 3 are shown in Table 1.

Table 1. Test parameters for CUR test circuit

Charge voltage

Peak thyratron current

Thyratron RIH5 current

Pulse repetition rate

Pulse width

50KU

5KA

36KA

2D00pps

150ns

Sample output signals of the standard thyratron CUR are shown in

Figure 4A and Figure 4B.

THE PEAK IS 1999.72

Figure 4A

220
AMPS see

450

400

350

300

250

200

150

100

Figure 4B.

The risetime of the standard CUR in a 50 ohm circuit iiiith a mercury

switch is equal to or less than 1ns. It is important to note that the

noise free signal of the CUR at the low current level shown in Figure 4B.

Figure 4A is the thyratron current switching a 7.5nf capacitor at 25KU.

The discharge circuit is almost critically clamped to simulate the conditions

at which the capacitor will be used in a laser application. (Huch lower

current level/ 450 amp peak/ is shown in Figure 4B as the thyratron

current from a discharge of a 500pf capacitor at 25KU. The risetime of

the waveform is 35ns which is also the switching time of the HU5322 in this

particular circuit.

Response of the low inductive version of these thyratron CUR's has

been measured in a 50 ohm mercury switched circuit to be equal to or less

than 300ps. This particular design provides an extremely low noise output.

In fact, the output signal is noise free over three orders of magnitude from

10 amps to 1000 amps peak. Even though the CUR has not been tested at a 10KA

level, there is no reason why it should not perform with comparable quality.

Two (2) of these CUR's are presently in use in parallel thyratron

arrangement seen in Figure 5.

The output signal of the low inductance CUR at a pulsed current of

15 amps is shown in Figure 6. The risetime of the current pulse in Figure

6A is inductance limited by the PFN. The output is noise free and the

measured pulse current matches the calculated current exactly.

221
Filgure 5,

Figure 6.

MIPS

THE PEAK IS 54.2664

Filgure 6A.

222
4. Stripline CUR - UJ Series

The stripline type of energy storage netiuork has become very common

due to the faster risetime and shorter pulse requirements of many lasers.

The stripline type CUR (Series UJ) is the best method of monitoring the pulse

currents in these fast networks and is shown in Figure 7. This type CUR haue

risetimes of 1 ns or less. These stripline CUR's range from a few to several

feet in width and can be mounted directly in the current return on laser heads

of considerable width. Circuits using plate conductors (to minimize inductanc

similar to the one shown in Figure 8 # are common applications of this flat CUR

Output signal of the flat CUR is shown in Figure 9.

Figure 7.

Figure 8.

223
IE 3 AMPS

1 THE PEAK IS 1043 22

-.2

1E-9 S

Figure 9.

This waveform is the discharge current pulse of a 7.5nf capacitor at

25KU. It should be noted that this is the same current pulse as is shown in

Figure 4A and that was measured with the thyratron CUR. Since the thyratron

CUR and the flat CUR were used in the same loop as seen in Figure 3, the

currents should be identical. Superimposing the output signals of the load

CUR and the thyratron CUR in the discharge loop of a 500pf capacitor at 25KU

we get the waveforms shown in Figure 10.

The flat CUR's have a measured bandwidth of DC to 300tllHz in a 50 ohm

circuit with a mercury switch. Figure 6A shows the risetime of the mercury

pulser and is less than 1ns and the risetilme of the CUR to be faster than

the PFN. .

AMPS

500

450

400

350

300

250

200

150

10O

20—40 €0 80 100 12 1E-9 S 140 160 ISO 200

Figure 10

224
5. di/dt Probes

fllany experimental systems where isolation requirements may require the use of wrap around Rogowski coils. Two specif ilea tions are generally required

for probe selection, they are sensitivity and bandpass. Keeping within the

operational frequency range, the probe output voltage is given by the formula:

Output voltage

Amps per mic rosecond (di/dt)

Probe's se nsitivity (5

)

(3)

The probe's sensitivity is depentent upon the probe's mutual inductance,

resistance of the probe, and load impedance as shown below:

Sensitivity (S) Impedance(Z) + Resxsta nee (ohms) •

, — . . , . c ; 7—t s x [Hutual I nductance (ITIJ (4j

I mpedance \Z)

(Hutual I nduc ta nc e ( (11) = I nductance (nh) x I (Amps per microsecond) (5)

Bandpass can be similarly shown to be primarily limited by the potential

drop across the probe's self-inductance. Neglecting resonance effects the

bandpass (3db down point) is given by:

Bandwidth (Hz)

Resista nee (ohms) + lmpedance(z)

I nductance (L)

(6)

However, it should be noted that probe oscillations may dominate over

this simple relationship.

In' the development of many pulsed power devices, "risetime" and "rate

of rise" of currents becomes important quantities. Direct measurement of

risetime was simplified with the use of computers and waveform digitizers

and the rate of rise is measured directly by di/dt probes like the one shown in Figure 11A.

Figure 11A

225
The dil/dt probe is placed around the conductor of the discharge loop.

Ulhen used in high peak power pulsed circuits, insertion is negligible.

Another type of probe is simply placed on a conducting surface. This

type probe, shown Figure 11B, is particularly useful in stripline circuits.

An -important feature of this type probe is that it does not haue to encircle

the conductor. This is especially advantagous in extremely high voltage

circuits. Both types of these flat di/dt probes must be terminated into

50 ohms and then read by the test equipment or integrated by a passive

integrator.

Figure 11B.

Figure 12A shows the output of the di/dt probe supplied with the CUR

for a Hy-5 Thyratron.

Figure 12B shows the output of the passiue integrator when fed by the

output of the di/dt probe.

Figure 12C shows the output of the CUR itself when attached to a HU-5

Thyratron.

This integration of the di/dt signal gives a true current waveform output.

Ulhen computor controlled digitizers are being used then the probe output

can be integrated by a computor program. A sample program using a Tektronic

Word Processor Transient Digitizer is shown in Appendix A. This program prints

a graph of the di/dt signal, maximum di/dt rate, and a graph of the integrated

di/dt signal.

226
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Figure 12C.

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230
Appendix "A" Graphs

Actual Output

THE S NUMBER IS 8245.54

THE DI/OT IS 1.83618E+18

Real time plot

of d-u/dt

AMP

I ntegra ted

Output

40«

300

290

lee

-188

231
References

Park, John H., "Shunts and Inductors for Surge-Current measurements

Journal of Research of the National Bureau of Standards, Uolume 39,

September 1947.

232
WIDE-FREQUENCY-RANGE CURRENT TRANSFORMERS

AND APPLICATION TO PULSED POWER SYSTEMS

John M . Anderson

Electrical Systems and Technology Laboratory

Electronic Power Systems Branch

Corporate Research & Development

General Electric Company

Schenectady, New York 12301

Current transformers, constructed as a uniform toroid,

can have excellent high frequency response into the microwave region. Their natural isolation against ground loops

and ability to feed standard 50 ohm coaxial lines make them

ideal for current sensing in high-power pulsed systems.

Transformers having response from 4xl0~2 Hz to ^300 MHz,

ixt product of 72 Amp-sec (250 kA for 290 us) and droop of

0.025%/ms have been constructed with all-ferrite cores.

These transformers have been applied to measurement

of current near current-zero in the testing of high-power

vacuum interrupters. Currents ^10 kA were commutated to

zero in 10 us and post arc currents lasting 2 ys with 500

amperes peak were detected with current transformers. The

ability to introduce opposing currents in the window of the

transformer allows cancellation of capacitive current

driven by the large recovery voltage.

Key Words: Current transformer; wide-band transformer

1. Introduction

Properly-constructed current transformers (CT's) can

be made to have better frequency response, load the primary

circuit less, and give greater output voltage than shunts.

Better high frequency response is possible because the value

of burden resistance can be larger than the equivalent shunt

and therefore suffer less from skin depth or self inductance. Also the burden can by physically smaller because

it absorbs less continuous watts or impulse Joules. Resis233
tance reflected into the primary circuit may be less because

it is transferred according to the turns ratio squared.

Current transformers do have a low frequency roll-off, although this may be made 3 dB down at a very small fraction

of a Hertz.

In previous work [1] we have described the construction

of small CT's which showed uniform response to several

Gigahertz. However, their small size precludes use in high

power pulse circuits. A somewhat larger CT with pass band

from less than one Hertz to about one Gigahertz and ixt

product of 7 ampere-seconds has found general use at our

laboratory. More recently a larger transformer, also with a ferrite core, has been used for transient measurements where

its ixt product of 72 ampere-seconds with a 0.01 ohm burden

made it more applicable for high current pulses. Presetting the core flux so that AB can be at least 4000 Gauss

permits observation of a half cycle at 60 Hz up to 13.5 kA

peak. A window diameter of 6.3 cm permits use at very high

voltages when properly insulated.

In the following a very brief description of wide frequency range CT's will be followed by more specific construction details. The application of this transformer to

observe post arc currents in high power vacuum interrupters

will be described to show at least one case where its unique

properties are helpful.

2. General Discussion

A toroidal secondary winding is excited uniformly by

the magnetic field from a single turn primary if the primary

wire is coaxial with the principal torus axis. Independence

from circumferential variation means that the current is

induced to flow with near equal phase and amplitude at all

parts of the toroidal winding, and sensing this current by a small-value resistance gives an overall time delay of

response which approaches that imposed by the retarded potential. This structure is thus capable of a very rapid

rise time or high frequency response.

Figure 1 shows the elementary physical structure and the

secondary current which flows in response to the primary current. For a moment one can imagine the secondary to be a continuous conducting box in toroidal shape. The surface

current which flows is exactly that necessary to exclude the

magnetic field from the interior of the box, and its integrated magnitude is simply ip ampere turns, where ip is the

current in the primary wire. If the box is slit into n s helical turns the current in each turn is ip/n s in agreement

with ideal transformer conditions. A conducting shield with

appropriate slit around the toroid has currents induced on

234
its surface which serve to reproduce the external magnetic

field in the vicinity of the toroidal secondary as if the

shield was not present. The above is true only on a transient and quasistatic basis. Current in the secondary

is sensed by interrupting the winding and connecting, by a ti ght 1 y- twi s ted pair or coaxial cable, a burden resistor

outside the shield.

tip

Figure 1. Cut-away sketch of current

transformer primary and secondary orientation

The transformer described so far should show little

dispersion and should have a response delay of only a few

tens of picoseconds, even for a large structure 30 or so centimeters in extent. Where, then, are the limits to

frequency response? They are primarily uniformity of

structure and internal resonances. The first prominent

resonance in the few megahertz region is a coil resonance

where the coil inductance resonates with stray capacity to

core and shield. Characteristic of this resonance is a variation of fields azimuthally on the toroidal winding.

We have found [1] that a network of resistances around the

secondary as shown on Figure 2 damps these spurious responses without affecting the desired signal current. The

next resonance of consequence is due to the shield and

235
secondary acting as a re-entrant cavity. This can be damped

by placing a low value of resistance uniformly across the

slit in the shield. The correct value, generally 1 to 3

ohms, critically damps the resonance before eventual capacitive roll-off at higher frequencies. Feeding the CT from a coaxial cable where the outer conductor is interrupted by

the slit in the shield removes currents from the outside of

the shield and extends the frequency response as well as assures uniform azimuthal excitation, but application at high

voltage points in a non-coaxial circuit is precluded. A

notable exception might be a coaxial high voltage transmission line as used in gas- i nsul ated substations.

Figure 2. Arrangement of resonance-damping

resistances placed around the secondary winding

(taken from Ref . [1 ]

)

236
Mid-band and low-frequency responses are straightforward and easily cal cul ated . [1 ] The magnetic core acts to

maintain a gradient of tangential H across the secondary

until it saturates, maintaining a "surface" current in the

secondary, which is the desired signal. At very high frequencies the core material does not respond magnetically and

approaches a dielectric, possibly lossy. Transition between

low and high frequencies is smooth with constant CT sensitivity. At very low frequency, flux buildup and decay in the

core leads to a phase shift of output voltage for a.c. primary currents, 45 degrees at the 3 dB point^ and -to a small

negative oyerswinq following a Drimarv current Dulse, ?. . 3 mV it the full lxt product is used (0.01 ohm burden).

3. Current Transformer Construction and Testing

The transformer secondary should be in close proximity

to the core and wound with a wire size which leaves little

space between turns at the inner radius. We have used round

magnet wire, 10 gauge, although one might argue that rectangular wire would be better. Because the CT is a very low

impedance device, there are no places where large voltages

develop and insulation need be only for a few hundred volts.

The inside of the shield must be a uniform distance from the

winding to keep a constant magnitude of magnetic field at

the winding surface. Wires from the secondary are tightly

twisted and go immediately into a small channel to the outside of the shield to minimize induced voltage. The burden

is attached coaxially for best frequency response.

The density of damping resistors around the secondary

can be increased by experiment until coil resonances disappear. We have found that 1.3 kilohm 1 ow- i nductance film

resistors attached every 4 turns on a 100 turn secondary

suffice. They are commonly tied to a thin copper washer

which is insulated from the windings and the shield as shown

on Figure 3. If the transformer is worked directly into a terminated 50 ohm cable, i.e. a 50 ohm burden, the loading

from these resistors may reduce the output voltage at midband a few percent. Generally the burden for pulsed high

current observation is 0.1 ohm or less and loading is negligible. Connection to the oscilloscope is by 50 ohm terminated coaxial line which alters the burden resistance by

a very small and known amount.

The ferrite core was obtained by special order from

Ceramic Magnetics Company and had dimensions O.D. 17.4 cm,

I.D. 9.5 cm, with thickness 2.8 cm. The material was MN60

which has an initial permeability of 6500 and saturation

flux of 4500 Gauss. Remanent flux is cataloged at 1200

Gauss. Overall dimensions of the CT shown on Figure 3 are

20.3 cm square by 4.76 cm thick. Internal cavity dimensions

237
Figure 3. CT disassembled to show

construction

of O.D. 19.4 cm, I.D. 6.67 cm and height 4.13 cm allowed

ample space for the core and winding. Burdens of 0.1 and

0.01 ohm were obtained from T and M Research Products.

Catalog numbers were SBNC-1-1 (700 MHz, 40 Joules) and

SBNC-2-01 (300 MHz, 16 Joules), respectively. Their d.c.

resistances were spec'd to within 1%, but values are of

course measurable as received to greater accuracy. We have

observed no internal resonances, and rise times for the burdens alone are in reasonable agreement with the cut-off

frequencies given above. Some calculated performance data

for the CT described above are given on Table 1.

4. Measurement of CT Response

Mid-band sensitivity (transfer resistance) of the current transformer can be rather simply tested in the audio

frequency range. We have used a General Radio Type 1450 TB

Precision Decade Attenuator followed by a Hewlett-Packard

400F voltmeter as a constant voltage indicator only. A

resistance, generally one ohm, was alternately used as a burden or directly fed from the same constant current source

used to feed the CT. The difference in attenuator settings

as it was connected to measure the voltage across the resistance in the two cases gave the current ratio of the CT.

Precision of visual reading was well within one percent.

238
Table 1 Load (Burden) Resistance 0.1 0.01 ohm

Voltage, out for 250 kA 250 25 volt

primary current

ixt product for AB=4000 Gauss* 29.3 72.5 Amp-sec

Pulse length at 250 kA for 117 290 ys

ixt listed

Impulse energy in the burden 73.1 18.3 Joules

for above pul se Secondary inductance measured 239 239 mH

at 1 kHz

L.F. cutoff* (small signal) 0.098 0.038 Hz

Droop* (small signal) 0.062 0.024 %/mS

Resistance reflected 1.5x10-5 6.1 x 1 0

6 ohm

to primary*

*Includes secondary winding resistance of 0.051 ohm.

The attenuator was limited by internal resistor accuracy of

±0.25% (±0.02 dB ±.25% of indicated dB). By 200 kHz this

possible error increases to 0.1 db ±1%. At 400 Hz and 100

kHz the turns ratio of the CT agreed with attenuator readings so that much less than 1% voltage change was seen.

This result should be expected if the permeability of the

core is adequate and losses are small. CT accuracy then

depends principally upon the burden which can fortunately

be wide band and accurately measured at d.c.

The CT, unless damaged, is mechanically and electrically stable, which leaves oscilloscope techniques and

physical placement in the circuit as the most probable

sources of inaccuracy and non-reproduci bi 1 i ty . For the

most uniform response at frequencies above about 10 MHz,

the primary wire should be centrally placed in the CT

window. This reduces, non-uniform excitation of the secondary. Where coaxial excitation is possible, the requirement

is immediately met. In other cases the primary wire should

be coaxial for 3 to 4 diameters of the secondary to minimize

error, although this was not quantified.

Testing the transformer for highest frequency response

is most easily done by a fast step of current. To generate

the step, we have used an air line, discharged through

mercury-wetted contacts. The current is limited by the

contacts to ^one-half ampere, that is a 50 volt line dis239
charged into 50 ohm. The current step so obtained is shown

as the top trace on Figure 4. Its rise time is ^200 ps.

The output of the CT with 10 ohm burden is shown as the

lower trace. At first glance the response looks to be a disaster, but it is mostly explainable and correctable for

our purposes. Noise on the trace is due to a high gain

setting on the sampling oscilloscope. The oscillations at

2 GHz are largely cavity resonances and were not fully

damped by 3.2 ohn placed across the slit in the shield.

They could have been further reduced but were not. The

possibility of some spurious responses due to winding irregularities should not be discounted. The CT response does

not step to the same relative amplitude as the current source

and, in addition, rises more slowly thereafter. This is

caused by a too-large burden, 10 ohm, necessary because of

equipment limitations, and tests on other transformers [1]

shows that the effect will reduce proportionately when the

burdens of 0.1 and 0.01 ohm are used.

CURRENT

SOURCE

CT RESPONSE

TIME

Figure 4. Response, lower trace, of the CT

to a step current, upper trace, showing residual

cavity resonances.

We expect to use this CT with oscilloscopes responsive

to ^100 MHz and therefore a 100 MHz roll-off was imposed on

the source step current, see the top trace on Figure 5.

The CT response is shown on the lower trace. Rise time with

the 100 MHz roll-off was 3.5 ns and with the CT in addition

it was ^4 ns. Noise on the trace does not permit accurate

measurements. However, using the approximation that the

240
cascaded rise time is the root of the sum of the squares of

individual rise times, the CT in responding to a cut-off

^170 MHz. This will improve if a lower resistance burden

is used as noted above.

CURRENT

SOURCE

CT RESPONSE

TIME

pqp BBS

mamm

m mmm

a 1

MP n

WA m n | _

5 ns

by a

Figure 5. Response,

a current step with high

L-R rol 1 -off at 1 00 MHz .

lower trace of the CT to

frequency response limited

Applications

In the past few years we have used CT's of this design

in various sizes for a variety of applications. Coax i ally

fed, they have been used for example to observe sub-nanosecond-rise-time pulses of current to light emitting diodes,

current pulses in plasma display panels, and microwave currents to antennas. Larger size transformers match nicely

in frequency response to laboratory oscilloscopes good to

>1 00 MHz. The CT described in this report was used to

observe post arc currents in vacuum interrupters during

synthetic testing of their interrupting capabilities. [2]

A large capacitor bank was switched through an air inductor

to reproduce a one-half cycle current pulse at 60 Hz. A

second, lower capacity bank, charged to 100 kV, was switched

to create a counter-pulse in the vacuum interrupter and

bring net current to zero in ^10 us. If the vacuum interrupter does in fact interrupt at current-zero, it faces up

to 200 kV of reverse "recovery" voltage and residual plasma

241
is swept from the vacuum discharge space. Figure 6 shows

the post arc current, in this case about 200 amperes.

Recovery voltage was here about 150 kV. Time scale was one

microsecond per cm on the scope face. The available frequency response of the CT is helpful to observe noise on the

post arc current and exact times of failure to interrupt as well as the nature of the failure. Reference [2] gives

examples of failure which will not be reproduced here. In

addition to a divorce from direct connection to the circuit

necessary with a shunt, the window of the CT is available

to pass bucking currents for cancellation of stray capacitive

currents to or compare currents differentially in different

parts of the circuit.

500A/cm

CURRENT

VOLTAGE

46.5 kV/cm

TIME

Figure 6. Waveform of current near currentzero, upper trace, and recovery voltage during

synthetic testing of a vacuum interrupter. (Taken

from Ref. [2])

6. Ac knowl edment

s

It is a pleasure to acknowledge the assistance of

CP. Goody in the construction and testing of these transformers. J.J. Carroll collaborated with the author in the

testing of vacuum interrupters and A.N. DeTommasi assisted

in set up and testing.

242
7. References

[1] J.M. Anderson, "Wide Frequency Range Current

Transformers", Rev. Sci. Instrum., 4_2_, pp. 915-26,

(1971 ).

[2] J.M. Anderson and J.J. Carroll, "Applicability of a Vacuum Interrupter as the Basic Switch Element in HVDC

Breakers", IEEE Trans. PAS, PAS-97 , pp. 1893-1900,

( 1 978 ) .

8. Sponsor Acknowledgment

Testing of vacuum interrupters was supported by a contract with the Energy Research and Development

Administration, now the Department of Energy.

243
Discussion Session 4

Richard Malewski (IREQ). My experiences with current measurements

were different. There are several common points in measuring techniques

used by physicists who are here and power engineers. I feel that

probably power engineers know more about shunts whereas the physicists

involved in the measurements of very fast pulses know more about

current transformers or Rogowski coils. I believe that we have made

a contribution in the area of shunts. Shunts must meet the two

contradictory conditions. One is to provide sufficiently large volume

to absorb the energy dissipated during the measurement of current

pulse, while the second is to have sufficiently fast response to

reproduce the measurement pulse. Unfortunately, the voluminous shunt

has fairly thick walls and, therefore, the skin effect slows down the

response. The classical tubular shunt such as devised by Francis Silsbee

is a tubular conductor having certain thickness of the tube wall, If

the shunt is subjected to a unit step of current, the current density

initially is very high at the outer wall of the shunt and is practically

zero at certain depths. The current density will gradually penetrate

into the shunt until a steady state is obtained. If our measurement

circuit is attached to the inner wall of the shunt, as proposed by

Silsbee and Park, the voltage which builds up will have a shape

representing the gradual increase of current density of the inner

wall of the shunt. The increase of the response is dictated by the

rate of penetration of current density into the material. We would

like to shorten the response of the shunt. This can be done in two

ways. The first is to take the probe and introduce it inside the

wall of the shunt at a given radius. There is an optimum position

between the inner and outer wall for the probe. Technologically, one can laminate this shunt out of two tubes and introduce this probe in

between. The second other improvement is technologically more complicated. Assume that we cut the shunt tube with a longitudinal

cut. By introducing the probe into the shunt wall and running it

along a parabolic profile, the response will be an exact replica of

the applied pulse. Therefore, we do eliminate error caused by skin

effect in the shunt and also conserve the large volume of the material

necessary to prevent thermal drift of resistors due to heating. This

shunt has been described in the literature J This shunt could be

incorporated into some of the systems I have seen here shown during

the discussion. In the area of current transformers, I have a question

which I would like to address mainly to two authors who presented

their design of this transformer. If I am right, the main problem

R. Malewski, et al . "Elimination of Skin Effect Error in Heavy Current

Shunts." IEEE Transactions, Vol. PAS-100, No. 3, March 1981,

p. 1333-1 340.

244
with current transformers comes from non-axial excitation. If we assume that the current flows through a conductor which is put

asymmetrically into a current transformer we see immediately that the

magnetic field generated by this current will be symmetrical along

the axis of the conductor. This means that different parts of the

coil will see very different field intensities. The current induced

into secondary winding around the donut will have a tendency to cancel

the primary flux but the current recirculating in this winding will

produce a flux of opposite direction but of the same value all around

the coil. Under transient conditions we will have some net flux

inside the coil which results because the primary flux is stronger

than the secondary at some locations (figure 1). Such a situation

cannot exist for a long time and energy stored in the magnetic field

of this coil will be dissipated showing up as ringing due to some stray capacitance and inductance in the coil. This is the reason why

people designing these coils added a burden in the form of distributed

resistance attached to some sections of the coil. In this way,

current flows in the small local loops in addition to the total

currents circulating in the coil (figure 2). And this will allow the

flux due to the current in the coil to bond better, perhaps stepwise

the primary flux. I once opened a transformer made by Pearson and

found inside the transformer a winding made in such a way that there were taps from the coil bonded to a resistive wire with the total

resistance corresponding to the burden resistance here. There were

also some parallel high ohmic resistors probably for damping resonances

in the winding. Each of them was in the range of 500 ohms. The

resistive wire was, I think, about 50 ohms. There was also an apparently controlled capacitance of the winding elements to the

grounding refe rence slab in order to build the line of a known

impedance Z = /L/C. I believe that by changing the inductance by

using for instance tape or ribbon, instead of wire, and controlling

this capacitance one can reduce the impedance of the line and,

therefore, this would project on values of the resistors. I don't know

exactly how the transformer works as we have in parallel a fraction

of an ohm resistive wire section and several ohms of the radius-type

resistors, and the two somehow perform their roles despite the obvious

fact that one seems to be short circuited by the other. Perhaps

Mr. Pearson could explain this if it is not infringing with the

proprietary information of the company.

245
Figure 1. Schematic representation of flux in a current

transformer with off-axis excitation.

246
Figure 2. Schematic representation of flux in a current

transformer with off-axis excitation and distributed

load for i 2-

247
DATA ACQUISITION AND PROCESSING TECHNIQUES

R. E. Kolibas

P. A. Corbiere

J. J. Moriarty

Raytheon Company

Missile Systems Division

Bedford, Mass. 01730

ABSTRACT

Current measurements, signal transmission, and system fault

isolation of a line type modulator have been accomplished in the

presence of pulse power noise. This paper presents techniques

successfully utilized to achieve valid results.

1. Introduction

This paper describes techniques of data acquisition and processing

in the high noise environment of a pulse power system. These techniques, which are considered to be generally valid, are illustrated by

means of an example taken from the design of a hardware system which

has been built, tested and delivered.

The objective in this example was to monitor the operation of an

array of high power thyratrons and to provide fault detection, fault

isolation and system shutdown commands to a high- voltage, line -type

modulation. Manual test procedures by trained personnel were

unacceptable in this system both because of the hazards involved and

the large number of thyratrons to be monitored.

The parameter monitored and measured to determine valid operation was the thyratron cathode current. Both the timing and the amplitude of the thyratron current were analyzed to determine that all

switches fired concurrently within the limits required for proper operation of the line-type modulator. In the event of a thyratron malfunction,

the detection, processing and storage of the failure information as well

as the generation of a system shutdown signal, under predetermined

fault conditions, were provided by the thyratron monitor.

248
The key problem was to achieve valid operation despite the large

electromagnetic radiation, electrostatic coupling, and ground current

transients. The following discussions will explain the analog noise

suppression techniques as well as the operation of the monitored and

transmitted thyratron cathode current signal. The isolation of large

ground voltage transients between subsystems will be addressed.

Finally, the diagnostic methods utilized to perform the more complex

tasks of fault isolation, including digital techniques that can achieve

noise suppression via judicious logic architecture combinations, will be

presented.

2. Measuring the Cathode Current

The operating principle of the Rogowski current transformer is the

use of a toroidal core transformer as the secondary winding enclosing a one-turn high-pulse current source. The pulse current-carrying conductor in this case is the plasma formed within a thyratron switching a peak current in the range of 1000 to 10, 000 amperes with a plate potential of 45 kV.

The design premise of such a transformer is based on its reflecting

zero insertion loss into the high pulse current circuit being monitored.

In practice, the secondary circuit impedance reflected into the single

turn primary is made to be from 0. 01 to 0. 10 percent of the source

impedance by means of matching output coaxial impedances of 50 to

100 ohms.

Thus the equivalent operating circuit of Figure 1 may be used for

transformer and signal circuit test and evaluation. By means of a

PULSE AMPLIFIER

R

s « R L

I 1

I = I t

vvv-i

TERMINATION

i 0 i

Figure 1. Pulse current transformer equivalent

ope rating /test circuit

249
three-place digital or analog visual comparator (oscilloscope), the test

and signal circuits may both be calibrated well within the maximum

desired accuracy of 1 percent.

From a design point of view the current transformer leakage

inductance is part of the equivalent impedance reflected into the primary

circuit and therefore will not attenuate output circuit risetime. Current

transformer core loss and magnetizing current must be maintained

sufficiently low in order to be within the transformer linearity (or

accuracy) requirements over the operating current range specified.

Response to fault current levels may be permitted to degrade, generally

to the point of avoiding core saturation effects, and is specified as the

transformer It capability, where I is the peak pulse current and t is the

equivalent square wave pulse duration.

Output pulse droop requirements are met by appropriate output

winding inductance. In cases where nickel-iron core materials are

required for unidirectional input pulse currents, an isolated dc bias

core reset current is continuously applied.

Output waveform fidelity is maintained by distributing the built-in

terminating resistance symmetrically around the transformer such that

an upper cutoff frequency of 10 MHz is readily obtained. In this case a

1 -microsecond pulse current" risetime is faithfully reproduced and the

internal waveform ringing is damped by the load resistor distribution

technique.

3. Protecting the Monitored Signal

Shielding the monitored signal against electromagnetic interference

at the pulse current transformer includes a variety of techniques as shown schematically in Figure 2.

First and foremost is the principle of using a uniformly wound

toroidal core for the transformer. As a consequence only the net axial

current or magnetic flux through the inner diameter will induce output

current, all others being cancelled by the core and coil geometry.

Secondly the monitored signal is enclosed in a Faraday cage which

does not carry signal current. In the transformer this takes the form

of the low impedance case structure electrically connected to the shell

of the output twinax coaxial cable connector. A gap in the case inner

circumference is used to pick up current and to prevent a shorted turn

around the core. This Faraday cage is electrically grounded at a point

remote from the high pulse currents being monitored.

Lastly an internal electrostatic shield covers the output winding in

the same manner as the case. This shield is electrically tied to the low

end of the unbalanced current transformer output winding to short

potential signals electrostatically induced in the winding. Additional

electrostatic and magnetic shields may be used to line the case as required.

250
r

n

i +

1 <

1

1

1

1

1 ——

4

>

! J

HIGH

LOW

FARADAY

CAGE

Figure 2. Pulse current transformer electrical schematic

4. Protecting the Transmitted Signal

A balanced line transformer is used to couple the unbalanced

signals through the balanced windings as shown in Figure 3. Here,

r

PULSE CURRENT

TRANSFORMER

L

SHIELDED

TWINAXIAL

CABLE

BALANCED LINE

TRANSFORMER

I

1

I I

v

I I FARADAY CAGE

<

Figure 3. Pulse current signal transmission

251
unwanted common mode signals are rejected. This configuration uses

a twinaxial, twisted-pair, shielded cable. Protection against low frequency electromagnetic fields is achieved two ways. First, the twisted

wire cancels the induced H fields, and secondly, the balanced line transformer common mode rejection cancels the induced signal remaining.

Protection against capacitive electrostatic coupling is achieved using a Faraday cage technique with a nonsignal-carrying shield. Lastly,

isolation from multiple subsystem grounds is accomplished by this

floating pulse current transformer and balanced line transformer

configuration, thus enabling the monitored signal to be effectively

referenced at the data processing section.

The balanced line transformer design used in this application to

couple unbalanced signals through the balanced windings has a 50 dB

common mode noise rejection between input and output signals. Proper

design for common mode noise rejection across the frequency passband

requires that the input and output winding halves be balanced within

several percent for all winding characteristics such as resistance,

leakage inductance, and distributed capacity. This balance is obtained

by fully symmetric coil geometry implemented with attention to

mechanical tolerances. "Winding resistances are sufficiently balanced

by using wire from the same mill run and wound on the same machine,

with similar winding tensions, speed, etc. The winding turns ratio is

1. 0 to 1.0 and is strictly maintained between winding halves for the

necessary balance, i. e. , the winding center tap must be exactly held.

Flexibility has been maintained in this design to permit any electrical hookup desired for case, shields, and windings. Thus multiple use

of this one design is not only possible but has proved effective depending

on the application. As shown in Figure 4, each transformer winding is

r n

r i r

i

i

I !

1

BALANCED LINE TRANSFORMER

Figure 4. Balanced line transformer

252
individually screened. Electrostatic shielding of 50 dB can be attained

by connecting these shields to the nonsignal- carrying Faraday cage

shown in the signal transmission schematic of Figure 3. Although

additional electrostatic and magnetic shields could have been effectively

employed in this transformer, it was deliberately located in a relatively

low noise environment such that they were not required in the present

example.

The transformer design has upper and lower cutoff frequencies

approximately a factor of 10 beyond those of the transmitted pulse, so that waveform fidelity is assured.

5. Processing the Data

The thyratron current cathode signals monitored and transmitted

are now processed to determine valid operation. This analog signal is

terminated into its characteristic impedance and measured for a minimum level using a comparator. This signal is now time-processed and

analyzed by the logic circuitry. The achievement of valid circuitry

operation despite the large electromagnetic, electrostatic, and ground

current transients is addressed in the following paragraphs.

Problems encountered designing with the logic remain. The

unavoidable noise, regardless of grounding philosophy, signal distribution and isolation techniques still requires the logic circuitry to have a respectable amount of noise immunity. Transistor-transistor logic

(T^L) with only a 400 mV noise margin is not an adequate choice.

However, a logic series with both adequate noise immunity and complex

large scale integration (LSI) capabilities is not available.

The decision to use two different specialty logic series in combination is the solution. A high noise immunity (HINIL) series with limited

logic capabilities but with noise immunity nine times greater than T^L

is used as the protective interface logic. Sandwiched inside is a buffered complementary metal oxide semiconductor (CMOS) logic series

which performs the decision and memory storage functions. These logic

series are operated at a 12 V level which is more than twice the

standard 5 V logic levels. This results in a CMOS logic noise immunity

six times greater than standard T^L. Thus the judicious combination

of these two logic architectures results in a logic system that survives

the electrostatic, electromagnetic, and ground transients with adequate

noise immunity to reliably perform even the most complex decision

functions.

Processing of the detected thyratron cathode current begins with the

timing and control logic (Figure 5) which develops a prefire zone, a valid fire zone, and a misfire zone. This information controls the time

sequenced gating logic which then compares in real-time the detected

output to these zones. When a failure occurs, the fault decision and

computation logic defines and isolates the type of problem (for example,

253
PROCESSING THE DATA

PROCESSED

ANALOG SIGNAL

PROCESSED

ANALOG SIGNAL

TIME SEQUENCED

GATING

LOGIC

FAULT

DECISION

&

COMPUTATION

LOGIC

FAULT

STORAGE

&

DISPLAY

LOGIC

TIMING AND CONTROL LOGIC

I

SUBSYSTEM INTERFACE

Figure 5. Data processing

thyratron number 14 prefired). This information is stored and

displayed. This is accomplished by having separate detectors for each

thyratron tube.

Furthermore, after each line modulator's output an additional scan

occurs which checks each thratron's output for a misfire failure. This

type of failure could be either a thyratron that did not fire or fired late.

Thus after additional fault encoding is performed this information and

the pulse number during which the problem occurred is stored in a random access memory (RAM). This enables one to detect and store

failures of multiple thyratrons on the same pulse or multiple failures

of the same thyratron on different pulses. Finally the system shutdown

control signal is generated during this scan sequence. When the failure

meets a predetermined set of conditions the logic automatically powers

down the system, preventing serious damage from occurring.

6. Conclusions

The capability of automatically powering down the system to a safe

level, especially during initial test and integration is significantly

useful. The failure data stored is retrievable for display on the front

panel. Therefore during troubleshooting, this data, properly analyzed,

can reduce the complexity and time required to isolate the line -type

modulator problem areas.

254
Thus the complex tasks of automatic fault detection, fault isolation,

data storage and system shutdown can be achieved through the judicious

use of analog and digital noise suppression techniques. The techniques

of data acquisition and processing described above have enhanced the

value of fault protection for the integration, operation, control and

maintenance of pulse power systems.

7. Acknowledgements

The authors wish to acknowledge the many contributions received

during the performance of the work described in this paper. Thanks

are due to Mr. J. R. Arsenault and Mr. G. K. Simcox for sharing their

technology and much appreciated experience.

255
A TECHNIQUE FOR MEASURING BEAM CURRENT DENSITY

IN THE ANTARES ELECTRON GUN

W. Robert Scarlett

University of California

Los Alamos Scientific Laboratory

Los Alamos, NM 87545

The Antares Electron Gun produces a radial beam of high energy

electrons for controlling the discharge in a large laser power

amplifier. The gun delivers a 5 ysec pulse of 450-550 kev electrons

with a current density of 50-100 mA/cm^. The 9.3 m^ beam is delivered

through 48 foil windows.

Because of the low electron flux, traditional means of making

high resolution measurements of the time integrated beam uniformity

have not proven very useful. We have developed a technique by which

Cerenkov light produced in a plastic radiator placed above a foil

window is recorded photographically and then measured with a densitometer. This data is then reduced to provide the integrated

current density.

The design, calibration and performance of this technique will

be presented. An extension of the technique to provide time resolved

measurements will also be discussed.

256
THOMSON SPECTROMETER MEASUREMENT OF HEAVY ION BEAMS

PRODUCED BY A PULSE POWERED PLASMA FOCUS DEVICE*

M. J. Rhee

Electrical Engineering Department

University of Maryland

College Park, Maryland 20742

A compact Thomson spectrometer is constructed for the purpose of

observing helium, nitrogen and argon ions accelerated in a pulse powered plasma focus. The spectrometer employs a single aperture system

and CR-39 plastic as a detector. The energy, charge state and mass of

the ions are determined by analyzing the Thomson parabolas. Also

methods of obtaining the energy spectrum and the energy-resolved emittance from the same Thomson parabolas are discussed.

Key Words: charge to mass ratio; energy resolved emittance; energy

spectrum; Thomson spectrometer.

1. Introduction

The Thomson spectrometer is a versatile device which has the ability to analyze simultaneously the energy distribution of ions having a wide range of charge to mass ratios. Since this spectrometer does not

have focusing property like many other types of spectrometers, a well

collimated beam is required for higher resolution. Accordingly, a high

sensitivity detector system is required because collimation reduces

beam intensity. Kuswa [1] and others [2] have reported successful use

of the Channeltron Electron Multiplier Array (CEMA) of high gain (~ 10^)

as a detector in Thomson spectrometer. This high gain detector permitted the use of highly collimated beam to increase resolution and re- cording of single shot event.

The nuclear track detector in general has 100% detection efficiency

i.e., individual ions can be recorded within certain range of energy for

given species. Recently the CR-39 plastic plate is known as an excellent material for track detector [3] and has been used as a detector in

a Thomson spectrometer [4, 5]. The main advantage of this detector is

not only its sensitivity which enable use of a highly collimated beam,

but also its spatial resolution of ion tracks. Since individual ion

tracks are a few microns in diameter, the accuracy of measurement of

257
ion deflections may be an order of magnitude higher than other systems,

thus one can build a relatively compact device with such a detector

without sacrificing its resolution.

The two aperture system is commonly used to produce a collimated

beam. In the case of an approximately point source or parrallel beam,

the single aperture system has the advantage of simplifying the structure and the operation of the system. The aperture plate(s) is com- monly placed in the upstream of the deflecting field region. It was

found, however, that placing the aperture plate in downstream has some advantage if a single aperture system is employed.. In the latter case,

the measured deflection includes only the portion of the ion path where

there are no fields and the path is straight whereas in the former case

the deflection also includes the portion of the ion path where the

electric and magnetic fields are present and the path is curved. Thus

fringing fields in the transition region introduce errors in the ion

deflection in former case but not the latter. Furthermore, for the same target deflection, the beam cross-section size is smaller for the

latter case because the ions spend less time after the aperture in the

latter case than the former, allowing less time for beam spreading due

to space charge and emittance effects.

The ion beam investigated using the Thomson spectrometer is produced by a new kind of plasma focus device which is powered by a H.V.

pulse as reported earlier [6]. It is apparent that since the ions are generated in tightly pinched plasma column, the ion beam is generated

from an almost perfect point source and good emittance is expected.

The following two sections describe ion beam source and the Thomson spectrometer. The remaining sections are concerned with experimental results, their analysis and discussions.

2. Ion Beam Source

A Marx generator charged Blumlein pulse system [7], which was designed for the injector to the University of Maryland Electron Ring

Figure 1. Schematic of experimental configuration.

258
Accelerator, was modified to produce a pulse with positive output polarity. The output is coupled to a small plasma gun which consists of an anode (inner electrode) of approximately 5 mm diameter metallic rod

and a planar cathode (outer electrode) with a 1.25 cm diameter hole on

axis as shown in Figure 1. The upstream side of the hole is plugged

with 1 cm long molybdenum or stainless steel rod which as a 6 mm diameter hole on axis. The downstream side is covered with 3 mm thick

stainless steel plate with a 3 mm diameter hole on axis. After passing through a flowmeter and a needle valve, various gases are continuously injected into the plenum in the cathode where its pressure is

monitored by a thermocouple gauge. The gas is diffused into the main

vacuum tank mainly through the upstream hole in the plug rsesulting in

formation of a volume of localized gas density in the anode-cathode

area. When a pulse arrives in the gun area, a heavy discharge current

flows through the gas which quickly pinches itself off due to the m=0

MHD instability. As a result of the pinching off of the current car- ring plasma column, a very high voltage is induced in the gap of the

broken plasma column. Thus the ions and the electrons in the plasma

column are accelerated to an energy equal to the product of the charge

stage and the induced voltage and are moving in the positive and negative axial directions, respectively. Hence the ions and electron ac- celerated by this induced voltage are expected to have a broad and con- tinuous spectrum. The magnitude of the induced voltage is highly transient, because the current carried by the earliest accelerated electrons

and ions acts to reduce that voltage.

A compact spectrometer as shown in Figure 2 is comprised of deflection sector, an aperture plate, and CR-39 detector plate.

3. Thomson Spectrometer

Figure 2. Thomson spectrometer.

259
The deflection sector consists of two soft ion pole pieces of 3 cm

x 3 cm cross section, which are held by two brass rings (not shown) to

maintain pole gap of 1 cm inside the 10 cm diameter aluminum vacuum

chamber. Placed outside the vacuum chamber is a permanent magnet which

produces a field of 2.6 kG in the gap. The electric field parallel to

the magnetic field is obtained by biasing a stainless steel electrode to

high voltage. This electrode is mounted on one of the magnet poles and

is separated from it by an insulating lucite plate. The aperture plate

with single or multi-pinholes is placed in the downstream of the deflection sector. The pinhole is made by using a sharpened needle to drill a hole through a 0.05 ~ 0.15 mm thick copper foil. The size of pinhole is

readily adjusted from 25 ym - 250 urn.

A CR-39 plate is placed 1 ~ 2 cm further downstream of the aperture

plate. If the region of electric and magnetic fields has a cross sec- tional area large enough, more than one parabola may be formed on any

given shot through the use of multiple apertures. For example, Figure 2 shows the use of a three pinhole system in which parabola at three different verticle positions are formed simultaneously. Because of high

spatial resolution of the detector, the size of the parabola track pattern can be as much as two orders of magnitude smaller than similar patterns on conventional types of detectors.

The deflection of ions in the Thompson spectrometer is well known;

ions of a given charge state and a distribution of energies appear as a parabola on the detector screen. Consider a charged particle passing

through and at a right angle to a region of uniform, parallel and horizontal magnetic and electric fields of length L. The particle experiences two different types of forces, one in vertical direction due to

the magnetic field and the other in horizontal direction due to electric

field. Since, as was stated before, the deflection paths are straight

lines, one may consider deflection angle (normalized deflection) rather

than absolute deflection. The calculation of the deflection angles are elementary under the assumptions of uniform fields and small deflection

angles. The magnetic deflection angle is given by

where 0b is the magnetic deflection angle in radians, Z is the charge

state of the ion, e is the electronic charge, B is the magnetic field

in Tesla, L the length of uniform field region, A is the ion mass number, M is the unit nucleon mass based on C 12

, and T is the kinetic en- ergy of the particle in Joules. Similarly the electric deflection angle is found to be

b /2AMT

_ ZeBL

(1)

0 =

e ZeEL

AMT (2)

where E is the uniform electric field in V/m.

260
* Combining Equations (1) and (2), eliminating kinetic energy T, we obtain the well-known equation of parabola as

Q

. ^MEL

e

2 _ 6 Ze(BL)

2 b

4. Experiment

The pulser system is operated at relatively low voltage producing

typical output of + 300 kV. This output voltage is monitored by two

probes, a CuSO^ resistive divider immersed in oil section and capacitive

divider mounted on the vacuum tank wall as shown in Figure 1 . Also the

current is monitored by a dl/dt probe in the vacuum tank. Three gases,

helium, nitrogen and argon, are used as the source of ions in this ex- periment. The gas flow is adjusted by a needle valve in the range of

3 ~ 10 cc/min. which gives pressure range at the plenum of 5 ~ 25 m Torr.

The anode is 5 mm diameter molybdenum rod and the cathode is a stainless

steel plug with a 6 mm diameter hole on axis. The ion beam is monitored

by a Faraday cup which is immersed in a transverse magnetic field (1 kG)

so as to suppress the possible secondary electron emission. The Thomson spectrometer is placed at the end of drift chamber, 100 cm from the

cathode. The aperture plate has three 65 um pinholes spaced 5 mm in the

vertical direction and is placed 10 mm upstream of the CR-39 target

plate. Three shots are made to complete the exposure of each detector

plate. One shot with both electric and magnetic fields forms the parabolas; the next with magnetic field only forms the vertical axis; and the

last with electric field only forms the horizontal axis. The exposed

CR-39 plate is then etched for 120 min. in 6.25 N NaOH solution at 70° C. Each plate shows three sets of parabolas corresponding r=0, r=+5mm

pinhole locations and the parabolas are found to be very similar to each

other. One of the parabolas for the nitrogen ion case is enlarged by

using a microscope and is shown in Figure 3(a). The term BL in Equations (1) and (3) is replaced by more realistic value / Bdl = 0.0133 TM;

i.e., numerical integration of measured magnetic field distribution

along the axis and approximate particle path. Unlike the magnetic field,

the electric field is very difficult to measure; the field of ideal parallel plate capacitor multiplied by the lengths of electrode is used in

place of the term EL in Equations (2) and (3). EL = (18 kV/cm) x 3 cm = 5.4 x 10^ V. With these measured values the Equation (3) is rewritten

as

= 3.164 f

0* , (4) e Z b

where the angles are measured in radians. From these equations the kinetic energy can be deduced from the parabola by measuring either electric deflection or magnetic deflection. A family of parabolas and possible values of charge to mass ratios for the nitrogen case are displaced in Figure 3(b).

261
Figure 3(a). Thomson parabola of nitrogen ions

enlarged from CR-39 detector plate.

0 100 200 300

9e (M RAD)

Figure 3(b). Calculated Thomson parabolas.

262
Also Equations (1) and (2) are rewritten with the measured constants

and in terms of convenient laboratory units for the kinetic energy

T

[MeV]

8.54 x 10

3

ao:2 (5)

b [m rad]

T

27 Z

(6) [MeV] 0e[m rad]

Despite the fact that the electric field in the spectrometer is approximated by an ideal parallel plate capacitor, the calculated and measured parabolas are in excellent agreement. The ions shown on the upper most parabola corresponds to charge to mass ratio of 7/14 for nitrogen which are fully stripped nitrogen ions, whereas the ions below

that parabola do not match the calculated ones for lower charge states

of nitrogen ions. Similar patterns of parabolas are found in the cases

of helium and argon gases as well. Comparison of these track patterns

with that of the theoretical curves for singly and fully ionized molybdenum ions supports the hypothesis that these tracks were made by various charge states of molybdenum ions, molybdenum being the material

making the anode. Since there are so many (42) charge states available

to molybdenum, the exact charge states of the detected molybdenum ions

could not be resolved with the present spectrometer.

As seen in Figure 3(a) virtually all of the ions from the gas are

in the fully stripped state. Also the energies of ions detected have

definite upper and lower limits. The lower value of energy is that

determined by the acceptance angle in the horizontal direction of the

spectrometer system. The Thomson parabolas for the other gases, helium and argon, were obtained under the same operating conditions as that for the nitrogen. The peak energies of those ions are easily

found by measuring either electric or magnetic deflections and using

Equations (5) and (6) . The results is plotted in Figure 4 as a function

of charge when the ions are fully stripped, i. e . , atomic number. It is

interesting to note that the peak energy is proportional to charge of

the ion, which is consistent with the prediction of the plasma diode

model [8] whose detailed description will be treated elsewhere [9].

An important characteristic of the pulsed ion source for its possible application is its energy spectrum. In principle one can con- struct an energy spectrum by counting the ion track density along a parabola, provided that the ion tracks are well separated from each other.

The track density along a given parabola as a function of either magnetic or electric deflection angle, which is a function of energy,

5. Results and Discussions

263
5 10 15 ATOMIC NUMBER Z

20

Figure 4. Peak energies of fully stripped helium nitrogen, and

argon ions as measured by the Thompson spectrometer.

can be found experimentally by using a optical microscope. Then the

energy spectrum is constructed by using the following relation

dN(T) dn d0

dT " d0 dT

(7)

where dn/dO is the measured track density in 0 direction, (either 0e or

Qfo direction) K is the ratio of total cross sectional area to the pinhole cross sectional area of the spectrometer. Unfortunately, the

track density of the parabola displayed in Figure 3(a) is too large for

individual tracks to be distinguished, thus not allowing the construction of an energy spectrum. A simple remedy for this problem is to

either use a smaller pinhole or move the detector plate further from

the aperture such that ion tracks are well separated.

Another application of such a Thomson spectrometer is an energy

resolved emittance measurement. Assuming a cylindrically symmetric

beam and no aberration, the normalized emittance is approximated [10] as

(8)

where a is the radius of the beam, r^ is the diverging angle, dr/dz at

r = 0, 3 = v/c, and y is ratio of relativistic to rest mass. The emittance of helium ion beam produced using this principle was estimated

and reported earlier [6],

264

e = 3yar' , n o
The same parabolas can be used to determine the energy resolved

emittance. Consider an idealized case in which the size of pinhole of

the spectrometer is inf initesimally small. Then a half of the line

width of parabola divided by the distance between pinhole and detector

plate is the diverging angle. In the case when the size of pinhole is

finite, the diverging angle may be approximately found by taking r' = \ (the full line width - dia. of pinhole) /separation distance. Thus

the energy resolved emittance can be calculated from the measured line

width along the parabola as a function of deflection angle which is

again a function of energy. The line width of the parabola for fully

stripped ions found in Figure 3(a) is shown to be nearly constant over

the entire length and measured to be 66 + um. Since the diameter of

the pinhole measured by a microscope is 65 + 1 um, the diverging angle

is too small to be analyzed for emittance calculation with reasonable

accuracy. Again smaller pinhole or larger separation distance or com- bination of both may improve the data reduction for such beam of low

emittance.

6. Acknowledgments

The author would like to thank L. Floyd for his assistance in the

experiment and for valuable discussions. J. Pyle also is to be thanked

for his technical assistance.

7. References

1. G. W. Kuswa, et al, IEEE Trans. Nuc . Sci. 20, 305 (1973).

2. J. N. Olsen, et al, J. Appl. Phys. 44, 2275 (1973).

3. W. Prior, et al, Bull. Am. Phys. Soc. 2A, 1039 (1979).

4. M. J. Rhee, Bull. Am. Phys. 25, 958 (1980).

5. F. Begay, Bull. Am. Phys. _25, 1010 (1980).

6. M. J. Rhee, Appl. Phys. Lett. _37 (10) , 906 (1980).

7. J. K. Burton, et al, IEEE Trans. Nuc. Sci. 2_0, No. 3, p. 321 (1973)

8. R. L. Gullickson, IEEE Conf. Record-Abstracts, 1980 IEEE

Conf. Plasma Sci. , p. 75.

, Int.

9. M. J. Rhee, to be published.

10. J. Fasolo, ERDA Summer Study of Heavy Ions for Inertial Fusion,

LBL-5543, 47 (1976).

8. Sponsor Acknowledgments

Work supported by NSF and AFOSR.

265
CURRENT MEASURING DIAGNOSTIC TECHNIQUES FOR

HIGH DI/DT PARTICLE BEAM ACCELERATOR*

D. H. McDaniel

Sandia National Laboratories

Albuquerque, NM 87185

Sandia National Laboratories has developed the 10 TW Proto-II

and 30 TW PBFA-I accelerator. The Proto-II accelerator has a dl/dt

of 2.5 x 10*4 A/s and has delivered 5 x 10^ A to an imploding plasma

load. PBFA-I has a dl/dt of 1.5 x 10 15 A/s and has delivered

1.5 x 106 A to an electron beam diode load. At these values of dl/dt

and current we have surpassed the limits at which current viewing

resistors (CVR's), Rogowski coils, cavity current monitors and B-dot

probes are capable of giving accurate measurements when closer than

1 5 cm to the 1 oad.

The accuracy of CVR's are affected when heating of the material

changes its resistivity during the current pulse. Rogowski coils are primarily limited 1) when the heating of the integrating resistor

causes it to change resistivity, and 2) when the voltage due to dl/dt across the coil housing causes breakdown. The cavity B-dot monitor

fails due to breakdown over the cavity opening at high dl/dt 's. B-dot probes fail when induced voltage causes breakdown of insulation

in which they are mounted.

The need to measure high currents at high dl/dt 's has led us to use piezoelectric gauges to measure the pressure induced in the

electrode walls by the currents in the accelerator. Situations in

which this technique is applicable are when the actual impedance at

the current monitoring position is much less than the geometric free

space impedance. This ensures that the electrostatic pressure effects

are negligible. These gauges have subnanosecond response when properly

mounted. The maximum pressure that these gauges can view, without

resorting to complicated code analysis of the signal, is presently

10*0 pascals. This corresponds to a level of 1.3 x 10^ A/m of liner

current density. The accuracy of the current measurements depends on the machining tolerances of the gauge and the accuracy at which

material shock impedances can be determined. This technique has been

used successfully on Sandia's Proto II accelerator and the Python

accelerator at Physics International. Piezoelectric gauges are currently being fielded on the PBFA-I accelerator at Sandia National

Laboratories.

*This work supported by the U. S. Department of Energy.

266
PROTON CURRENT MEASUREMENTS USING

THE PROMPT GAMMA RAY DIAGNOSTIC TECHNIQUE

R. J. Leeper, E. J. T. Burns, D. J. Johnson, & W. M. McMurtry

Particle Beam Fusion Research Department 4240

Sandia National Laboratories

Albuquerque, New Mexico 87185

Abstract

7 8 Prompt gamma ray signals from the nuclear reaction Li(p,"Y) Be have

been used to make time resolved proton current measurements. In these

measurements, the proton beam was allowed to strike cylindrical thick

lithium metal tarqets. The time inteqrated proton current was measured

using gamma activation of copper via the reaction Cu(Y,n) Cu(3+).

The positron activity of the copper sample was easily measured using

coincidence counting techniques. The number of Cu atoms produced per

proton incident on a thick Li metal target was determined with separate

calibration runs performed on the Sandia 2.5 MeV Van de Graaff accelerator. The time history of the prompt gamma production was measured using six EGG NPM-54 scintillator photowj 1 1 i p I ier combinations shielded

by 96.5 cm of concrete and 5.1 cm of Pb. The use of six scintillator

photomu I t i p I i er combinations was necessary to increase the statistical

precision of the data. The normalization of the prompt gamma time history data with the total time integrated proton current measurement

yielded the absolute time resolved proton current on target. Data from

runs performed on the Sandia Proto I accelerator will be presented.

1. Introduction

Intense, pulsed ion beams with current densities >_ 25 kA/cm have

been generated by reflex triode, reflex pinched and magnetically insulated ion diodes [ 1,2,31. The determination of the ion beam current

and current density for these diodes is difficult. In part, this is

due to the fact that intense ion beams are partially space-charge and

current neutralized. These traits preclude or make difficult many

electrical measurements involving charge collection or the measurement

of self-magnetic fields. In addition, the use of conventional nuclear

activation analysis [4] at these high current densities is precluded

because of beam induced target ablation. Furthermore, any screen used

to prevent target ablation will be vaporized and the holes in the screen thereby closed off. This subsequently prevents most of the

beam from reaching the activation sample.

267
In an effort to overcome many of these difficulties, this paper

describes a diagnostic method to measure the time resolved current of

intense proton beams by detecting the prompt gamma rays emitted from

the Li(p,Y) Be nuclear reaction induced in a lithium metal target

by the beam [51. The paper will also describe the procedure used to

calibrate the diagnostic. Finally, an experimental demonstration of

its use will be presented.

2. Diagnostic Method

When an intense proton beam pulse strikes a thick lithium metal

target, all protons above 440 keV in energy may induce the nuclear re- action Li(p,Y) Be. The prompt gamma rays emitted from this reaction

are produced in two energy groups of 14.7 - 15.2 MeV and 17.6 - 18.2

MeV. By measuring the time history of the prompt gamma ray production

with fast scintillator photomu I t i p I ier detectors, the proton current on target can be determined provided the \i(p,Y)^Be thick target cross- section is known. In principle, it is possible to calibrate the

scintillator photomu I t i p I i ers detectors directly on a Van de Graaff

accelerator. In practice, however, due to the intense Bremsstrah I ung

background associated with the ion beam production, the scintillator

photoml tip I iers require thick shields of some 76.2 cm - 96.5 cm of con- crete and 5.1 cm of lead for the preferential detection of the prompt

gamma rays. The use of these massive shields would require proton beam sources beyond the capabilities of conventional accelerators. In order

to achieve absolute calibration, we have used the photonuclear activation of copper via the reaction CuCY,n) Cu to measure the total number of prompt gamma rays produced. Then the proton current is

determined by normalizing the prompt gamma ray time history signal

with the total number of prompt gamma rays produced. A correction is

made for dependence of the thick target yield on proton energy.

The time history of the prompt gamma ray production required the use

of an array of six scintillator photomu I t i p I i ers to insure data of

good statistical quality because of the rather low \i(p,"Y)^Be production cross-section and the gamma ray attenuation in the massive

shield.

2.1 Prompt Gamma Ray Time History

The six detector array used to measure the prompt gamma ray time

history is shown in Figure 1. The scintillator photomu I t i p I i er combinations each consist of a 5.6 cm diameter x 6.7 cm thick cylinder of

NE— 111 plastic scintillator viewed by an Amperex XP2020 photomu! tipl ier

tube (6). The FWHM response of the detectors is 4.5 ns when used to

observe a single instantaneous event such as a cosmic ray. Each detector is capable of linear currents to 0.5 amp and is fully saturated at

1.0 amp of current. The detectors are operated at a sensitivity of 0.2

amps/(rad/sec) to Co gamma rays. The scintillators are coated with

white paint except for the photomu I t i p I i er tube viewing port. The

gains of each detector are cross calibrated using the Compton edge

technique. In addition, the gain of each detector is periodically

monitored for any shift throughout an experimental run. Typically

the gain shifts observed were less than \%,

268
WEIGHT = 3636 kg (8000 LBS)

108.7 em

Pb LID (5.1 cm THICK)

Figure I. Schematic of prompt gamma ray time history detector

The six detector array is required in present experiments to in- sure data of good statistical quality. It is found that on an individual shot (at an average signal level of 6 volts) there are variations

of + 80$ between individual detectors. By simply averaging the six

detectors together we produce a signal that is statistically better by

a factor of the yfE resulting in a net signal good to + 33$. It should be noted that a secondary benefit of a multiple detector array

is that it allows the statistical precision of the data to be measured

experimentally, whereas with a single detector, this is more difficult

to determine.

The massive concrete shield shown in Figure 1 is necessary to

attenuate the large x-ray bremsstrah I ung background that is generally

produced by electron loss in an ion diode. Whereas an ion diode may

be 80-90$ efficient in producing ions, there is an accompanying electron current which can amount to some 100 kA or more (10-20$ of the

total diode current) at voltages exceeding 1.5 MV. Experimentally,

it is found that 76.2 cm - 96.5 cm of concrete and 5.1 cm Pb are enough shielding to reduce this bremsstrah I ung background to less than

35$ of the prompt gamma ray signal for the I. 8 MV , I TW Proto I accelerator. The reason concrete is chosen as the primary shielding

material is because its gamma ray attenuation coefficient is a factor

2. I sma I ler at 15 MeV - 18 MeV than it is at 2.0 MeV [7.1. Consequently,

the concrete shielding will preferentially attenuate the lower energy

x-ray bremsstrah I ung background while passing the higher energy

prompt gamma rays with the overal I effect of increasing the prompt

gamma ray signal to noise ratio. The extension of this diagnostic

method to higher energy bremsstrah ! ung backgrounds , is not examined in

this paper and therefore use of this technique on other accelerators of

higher output voltages rrrus-t be careful fy investigated.

269
2.2 Total Prompt Gamma Ray Yield

A schematic of the experimental arrangement that is employed to

measure the total number of prompt gamma rays using photonuclear activation of copper is shown in Figure 2. The energetic gamma rays from

the Li(p,Y) Be reaction are allowed to strike a 7.6 cm diameter by

1.0 cm thick copper disk located 19.1 cm from the lithium target.

These gamma rays then induce the photonuclear reaction Cu(Y,n)

Cu(|3+). This reaction has on the average a 52.3 mb cross section

for the 14.7 - 15.2 MeV and 17.6 - 18.2 MeV 7 Li(p,Y) Be gamma rays,

and has a threshold of 10.9 MeV. The 10.2 cm thick polyethylene

moderator that encapsulates the copper disk is used to partially moderate and to attenuate any fast neutrons that might be produced in the

reaction Li(d,n) Be by the natural abundance of deuterium in the proton beam. This is important since neutrons above 11.0 MeV may produce

^Cu via the reaction ^Cu( n,2n)^Cu(8+) . This thickness of moderator

will attenuate these fast neutrons by a factor of approximately 18.5

and make their contribution to the activation negligible [8]. Note

that photonuclear neutron production in the moderator due to the

bremsstrah I ung background does not occur since the bremsstrah I ung end

point energy is below any photonuclear reaction threshold.

7Li (P, y ) 8 Be

63

67% 17.6 MeV GAMMA RAY

33% 14.7 MeV GAMMA RAY

62

Cu ( y ,n) Cu(/3+

)

PROTON BEAM

7.6 cm DIAM. X 0.95 cm

THICK COPPER DISK

10.2 cm

THICK POLYETHYLENE

NEUTRON MODERATOR

Li METAL TARGET

Figure 2. Experimental arrangement for measuring the total number of

prompt gamma rays using the photonuclear activation of copper

The Cu produced is a positron emitter with a 9.8 min half-life.

After activation, the copper disk is removed from the accelerator vac- uum chamber and placed between two 12.7 cm diameter by 7.6 cm thick

Nal (TZ) scintillator photomu I t i p I i er detectors. The two 0.511 MeV

annihilation gamma rays from the ^Cu decay are then energy selected

and counted in a standard coincidence counting arrangement. In this

way a highly sensitive and low background counting arrangement for

positron annihilation is realized. The background count rates are generally between 3 and 4 counts per minute.

270
3. Diagnostic Calibration

The absolute calibration of the copper disk was carried out on the

Sandia 2.5 MeV Van de Graaff accelerator. A thick lithium metal target

was irradiated by protons ranging in energy from 0.5 MeV to 1.5 MeV.

Target currents were generally 500 nA to 750 nA with target spot

sizes of 9 mm . Beam runs of 30 minutes were used throughout the

calibration. The 7.6 cm diameter by 1 cm thick copper disk was encapsulated in a 10.2 cm thick polyethyene moderator and located at an angle of 90° to the beam axis. After a beam run was completed, the

copper disk was removed and counted for a period of 20 minutes in the

coincidence counting arrangement described above.

Figure 3 shows the results of the calibration. Plotted are the

(background corrected) counts per gm Cu per incident proton on target

versus proton energy. The absolute detection efficiency of~5$ for

detecting positron decays from the copper disk has not been unfolded

from the data. It is clear from Figure 3 that the calibration factor

has a rather slow dependence on proton energy. The open circle data

shown is the thick lithium target yield data from Reference 9 normalized to our data at 0.9 MeV. The solid curve is obtained by connecting

the open circle data between 0.5 MeV and 1.1 MeV and then extrapolating

the curve through our data to 1.5 MeV. As can be readily seen, there

is good agreement between the present data and the data of Reference 9.

( 7.6 cm DIAM. x 0.95 cm THICK Cu DISK AT

14.9 cm IN 10.2 cm THICK MODERATOR)

4.0

ID

I O_ 3.0 o

za

2.0

1.0

0.0

—|

'

1

1 1 1 | ' I

' 1' 1 r

• SANDIA 2.5 MeV Van de Graaff Data (1980)

o L.R. Hafstad, N.P. Heydenburg, and M.A. Tuve,

Phys. Rev. 50, 504(1036)

(Normalized to SANDIA

Data at 0.9 MeV)

_i L

0.0 0.2 0.4 0.6 0.8 1.0 1.2 PROTON ENERGY (MeV)

1.4 1.6 1.8

Figure 3. Copper calibration factor Yqu versus proton energy

4. Experimental Demonstration

The current of a proton beam onto a 19 mm diameter by 12.7 mm

tall thick lithium metal target has been measured as an experimental

demonstration of this diagnostic method. The proton beam is generated by a radial magnetically insulated ion diode powered by the

271
Sandia Proto I generator. A schematic of this diode is shown in Figure

4. The copper activation sample is located 19.1 cm from diode center

and along the vertical axis of the diode. The six detector array is

positioned along the vertical axis at a distance of 1 .8 m from diode

center. A description of the operation of the diode is given in

Reference 3.

TO COPPER ACTIVATION SAMPLE AND SCINTILLATOR PHOTOMULTIPLIER COMBINATION ARRAY

TO PUFF GAS VALVE

Figure 4. Schematic of the Proto I radial magnetically insulated

ion diode

The larger of the two waveforms shown in Figure 5 is the signal

obtained upon averaging the signal from each detector of the six detector array for the case of the 19 rm> 4 i-ameter lithium target shot. The

copper disk activity on this shot was 128 counts/min measured starting

I 1 3 sec after the shot* The smaller waveform in Figure 5 shows the

average signal from an identical shot for the case of a 1 7 mm diam.

aluminum target. This signal is due to brems-strah I ung production from

electron loss in the anode cathode gap of the diode. The lower production cross section and lower energy gamma rays from proton induced

prompt gamma emission from A I combine to make their contribution to

the signal neglible MO, I II. The copper disk activity for this shot

was 8 counts/min at 130 sec after the shot. As can be seen, the peak

of the signal from the aluminum target comes earlier than the lithium

target signal and has an amplitude equal to 35% of the lithium target

signal. The delay- between the signals is due to t ime-of-f I ight of

the protons transversing the 6 cm distance from the anode to the

lithium target. Consequently the smaller aluminum target signal,

the delay between the aluminum and lithium signals, and the absence of

any counts above background on the copper disk for the case of the

aluminum target, when taken together, serves as positive evidence that we are indeed observing Li(p,"Y) Be prompt gamma rays.

272
-1.0

-7.0

'Li( P.7)' Be PROMPT GAMMA RAY AND

X-RAY BR E M MSST RAHLUNG BACKGROUND AVERAGE OF SIX DETECTOR ARRAY 19mm Li METAL TARGET

18 mm Al TARGET (BKG) PROTO I SHOT NO. 2935 and 2936

7 R Figure 5. Average Li(p,Y) Be prompt gamma ray signal and x-ray

bremsstrah I ung background signal

To determine the proton current from the normalization of the

measured prompt gamma time history and copper activation count, a correction must be made for the flight times of different energy

(voltage) protons from the diode to the lithium target. Then the

time variation of the prompt gamma ray detector voltage VQ

e+(+)

is related to the variation of the proton current I C + ) in ihe

f o I I ow i ng way

V Det (t) = K

J*

Y (V(t')) Ip(t') 6 (t - t' - i

v(t ) dt 1 (1

)

where K is a constant specified by the detector geometry, sensitivity,

and target-to-detector distance; V(t') is the inductively corrected

diode voltage at time t'; Y(V(t')) is a quantity proportional to the

thick lithium target yield for protons of energy V (number of prompt

gamma rays per proton); l D (t') is the proton current at time t' (number of protons per second;; i is the proton flight path from the anode

to the target and v is the velocity of protons of energy V = 1/2 Mpv^

emitted at time t' from the anode [12]. The calculated copper activation counts above background (C-B)p

g | is obtained by adapting to our case the standard activation formula of Reference 13. This relationship is

273
-\ti -Xto

(C - B) Cal

= M

Cu (e " e >

I <t«->6 (t-f \

\ v(t')/

dt'dt

(2)

Here M is the mass of the copper sample, X is the copper decay constant, t^ is the time interval between the end of irradiation to the

start of counting, ^2 ' s "*" ne time interval from the end of irradiation

to the end of counting, t

the time at the end of the voltage pulse,

copper activation yield curve given in Figure 3,

ables are the same as those defined for Eq.

is the time the voltage pulse starts, tp is

YCu (V(t»)>

(1)

s the absolute

and a I I other variThe first step in the data reduction routine we have adopted is to

determine the temporal shape of the proton current lp(t) from the prompt

gamma ray time history signal ^Defti")* ' n practice, this is determined

by iterating on Eq. (1) until the prompt gamma ray response calculated

with an assumed proton current agrees in temporal shape with the measured prompt gamma ray response. We use the absolute copper activation

yield YCu^ V

^'^ °^ F'9 ure 3 for Y(V(t')) since it is a quantity

directly proportional to the - lithium thick target yield. The initial

proton current assumed is the total diode current corrected for electron lost. Often this initial guess is sufficient to give good agreement between the measured and calculated prompt gamma responses. Once

the temporal shape of lp(t) has been determined, it is substituted into

Eq. (2) which is then integrated for the duration of the voltage pulse.

The net result of the calculation of Eq. (2) is the total number of

copper activation counts above background, (C-B)Q

g |, for the counting

interval t^ to t2 for the initial assumed proton current I (t). The

calculated copper activity (C-B>Q

a | is then normalized to The experimentally observed copper activity, (C-B)pX p, to yield a final proton

current given by

F i na I (t) = (C-B) Exp

(C-B) Cal

(3)

Applying the above data reduction procedure to the present data

yields the results presented in Figure 6. Shown in Figure 6a are the

final proton current and energy (diode voltage) used in Eq. (1) to

calculate the prompt gamma ray signal presented in Figure 6b. The

experimental signal in Figure 6b is obtained by subtracting the aluminum

target signal from the lithium target signal in Figure 5. In this way,

the Bremsstrah I ung contribution to the lithium signal is subtracted

away. The small prepulse in front of the large prompt gamma ray signal

in Figure 6b is due to this subtraction not being perfect. This prepulse being neglected, it is observed that there is excellent agreement

274
between the calculated and experimental prompt gamma ray pulses and that

there is 353 + 54 Ka peak current on the 19 mm diameter lithium target.

The error quoted is the addition in quadrature of a + 5.4$ statistical error from the copper activity used to obtain (C-B)p and an estimated

error of + 14.2$ in the value of the quantity (C-B>Q

a

j. This latter error is estimated from a calculation of (C~B)Q

a

|

' n which values of

YQ

u (V(t)) in Eq. (2) were allowed to deviate one standard deviation from

the mean. We note, however, that this rather modest error in (C-B)q , is due in part to the unique behavior of the voltage and current in this

experiment that just happens to emphasize the higher precision Yq (V(t')

data points of Figure 3. I n an experiment that has high currents at

substained low voltage this error will be somewhat larger. In yet

another parameter sensitivity calculation, it is found that varying

the voltage by + 10$ resulted in a variance of + 7.6$ in the quantity

(C-B)Q

a

|. No error in the quantity (C-B)q

9

| due to voltage variations

was included in the present error analysis. Finally, it should be

pointed out that the prompt gamma ray measurements only sample the proton current above 440 keV. Therefore, the measurement is insensitive

to the current profile for proton energies below this threshold.

TIME (nsec) Fig. 6a

Fig. 6b

Figure 6. The measured prompt gamma ray detector response (solid line)

in Figure 6b is compared with the response (open square

points) calculated using the proton voltage and current

displayed in Figure 6a.

5. Conclusions

A diagnostic technique for measuring the current of intense pulses

of protons with a time resolution of ~4.5 ns has been described. The

absolute calibration of a 7.6 cm diameter by 1.0 cm thick copper disk

used to measure total prompt gamma yield has been calibrated using the

Sandia 2.5 MeV Van de Graaff accelerator. Finally, as an experimental

demonstration of the technique, the absolute proton current onto a 19 mm diameter lithium metal target in a pulsed beam experiment was determined.

275
6. Acknowledgments

We wish to acknowledge the help and cooperation of R. A. Burchard

and W. R. Wampler in helping us perform the copper calibrations on the

Van de Graaff accelerator. Also we wish to thank J. Heise, 0. Marten,

and D. Tanner for their able operation of the Proto I accelerator.

7. References

[1] S. A. Goldstein, G. Cooperstein, R. Lee, D. Mosher, and S. J. Stephanakis, Phys. Rev. Lett. 40, 1504 (1978).

[21 J. Golden, C. A. Kapetanakis, S. J. Marsh, and S. J. Stephanakis,

Phys. Rev. Lett. 38, 130 (1977).

[31 D. J. Johnson, G. W. Kuswa, A. V. Farnsworth, Jr., J. P. Quintenz,

R. J. Leeper, E. J. T. Burns, and S. Humphries, Jr., Phys. Rev.

Lett. 42, 610 (1979).

[4] F. C. Young, J. Golden, and C. A. Kapetanakis, Rev. Sci. Instrum.

48, 432 (1977).

[51 J. Golden, R. A. Mahaffey, J. A. Pasour, F. C. Young, and C. A.

Kapetanakos, Rev. Sci. Instrum. 49, 1384 (1978).

[6] Available from EGG, Inc., Las Vegas, Nevada as Model NPM-54.

[7] R. E. Lapp and H. L. Andrews, Nuclear Radiation Physics , PrenticeHall, Inc., Englewood Cliffs, N.J., p. 424 (1972).

[81 S. S. Nargolwalla and E. P. Przyby I ow i cz , Activation Analysis with

Neutron Generators , John Wiley and Sons, New York, p. 30 (1973).

[9] L. R. Hafstad, N. P. Heydenburg, and M. A. Tuve, Phys. Rev. 50,

504 (1936).

[101 K. J. Brostrom and T. Huus, Phys. Rev. 7j_, 661 (1947).

[Ill G. P. Plain, R. G. Herb, C. M. Hudson, and R. E. Warren, Phys.

Rev. 57, 187 (1940).

[12] F. C. Young, F. Oliphant, and S. J. Stephanakis, NRL Memorandum

Report 4171 (1980).

[13] M. Barbier, Induced Radioactivity, John Wiley and Sons, New York,

pp. I3-I7 (\WTT.

8. Sponsor Acknowledgment

This work supported by the U.S. Department of Energy under contract

DEAC04-76-DP00789.

276
THE MEASUREMENTS OF PULSED ELECTRIC

CURRENTS BY THE FARADAY EFFECT

J. Katzenstein, W. Caton and G. M. Wilkinson

Maxwell Laboratories

8835 Balboa Avenue

San Diego, CA 92123

1. Introduction

The Faraday effect has been known for over one hundred years

and has been applied extensively to the measurement of magnetic

fields. Its use for the measurements of electric current is,

however, less common because the magnetic field arising from the

current flow in an arbitrary circuit in general has a spatial

variation. Hence, the calibration of a current probe employing the

Faraday effect involves geometric factors such as the dimensions of

the rotative medium and its position relative to the circuit [l] . An obvious way out of this difficulty is to design the optical path

in the rotative medium so that it encircles the current. Since the

Faraday rotation angle is proportional to the line integral of the

magnetic induction $ along the path i , we can write for a closed

1 ight path.

where V , the constant of proportionality, the so-called Verdet

constant, is characteristic of the rotative medium and has the

units of minutes of arc per gauss-centimeter.

From the circuital^ statement of Amperes law we may replace the

circuital integral $ B.d£ by 4tt/10 I where I is the current in

amperes that is enclosed by the path of integration. The resulting

Faraday rotation is thus

0 = V B« di (1)

277
The Faraday rotation produced in such a current probe depends

only on the Verdet constant of the rotative medium; geometric factors

do not enter. Current monitors based on this principle have been

developed [2,3], using optical fibers as the rotative medium.

Unfortunately, optical fibers which rigorously preserve the state

of polarization of light traversing them are presently beyond the

start of the art. As the Verdet constant of the fiber core material

(essentially fused quartz) is small, a large number of turns of the

fiber around the conductor is needed to obtain a measurable rotation

which greatly increases the stray birefringenece due to fiber strain

and curvature. To minimize the latter effect the radius of curvature is made large which increases the overall size of the probe.

Nevertheless, Faraday-effect current monitors have been developed

for use on high voltage utility transmission lines and offer a cost-effective alternative to conventional current transformers [4,5]

The light entering the Faraday probe is rendered plane polarized

by means of a polarizer (unless a laser emitting plane polarized

light is used as a source) and the light emerging passes through a second polarizer or analyzer before entering the detector. The

intensity of light & reaching the detector for an incident intensity of & is given by

where^ x "is the angle between the axes of transmission of the

polarizer and analyzer.

Two different methods of handling the Faraday effect signals

can be used depending upon whether the value of 0(1) at peak

current is very much greater or very must less than tt .

In the latter case, the usual practice is to make use of a differential analyzer such as a Wollaston prism which transmits two

output beams of intensity ^, and ^? which are polarized orthogonally to each other [4,5] . The analyzer is oriented at 45° with

respect to the transmission axis of the polarizer. The resulting

transmitted intensities are:

o

(3)

^ = ^ cos

2

(0 - 45°) , ^ = ^ cos

2

(0 + 45°) .

(4)

278
The sum and difference of these two quantities and their

quotient are computed. The resulting signal s is

J*, - ^

(5)

or for 0 « tt using Eq, 2

s = kl (6)

where k is a proportional ly constant determined by choice of

units

.

The resulting signal is then linearly proportional to I . The accuracy of the measurement depends on the accuracy which the

computations of Eq. 5 can be carried out. Since these involve the

determination of the small difference of two large quantities the

accuracy is critically dependent on the system signal-to-noise

ratio.

An alternative measurement technique useful when 0(1 ) >> tt is to use crossed analyzers (x = tt/2 ) . The signal is then

ax

^(t) = J

Q

sin^ 0(t)

1 - cos 20(t)

(7)

or using Eq. 2

1 - cos (?&vi(t))

(8)

The signal given by Eq. 8 can be inverted numerically to

obtain I as a function of t . Figure 1 shows the Faraday signal

resulting from a current waveform and its subsequent numerical

inversion to recover the original waveform.

279
1.20

1,00

0.80

0.60

0.40

0,20

Figure 1.

0 80 160 240 320 400 480 560 TIME (ns)

Sample current waveform and corresponding

Faraday rotation signal.

The accuracy of measurement is determined by the precision

with which the phase of cos 0 can be determined from the record

of I(t) which in turn depends upon the signal-to-noise ratio. If we call this phase uncertainty Acf> then the relative error £ is

5

=

ttN + 4> ttN (9)

where N is the total number of cycles of Faraday rotation. Thus, even though a relatively large value of A<j> is obtained due to a poor-signal-to-noise ratio, the error in the current measurement

can be made small by making N large. It is assumed, however,

that A<f> is not so large that a charge of sign of the time derivative of the phase cannot be detected. Such a reversal in d<J)/dt occurs when I attains an extremum.

280
In our particular application, the current I necessary to

change the phase of 0 by n is ^10 5 A. For the\urrents we wish

to measure N is 20-40. Thus, our maximum relative error is

2.5 percent. With a favorable signal-to-noise ratio this can be

improved by an order of magnitude (a resolution of 10 percent in

signal amplitude).

2. Detailed Design of a Faraday Current Monitor

for Large Pulse Currents

We have designed a Faraday-effect current monitor to measure

pulsed currents of the order of 4 MA max whose duration is typically

100 ns. The immediate application of the monitor is the calibration

of the magnetic pickup loops heretofore used for current measurements

on the BLACKJACK 5 pulse generator. The tests will initially be

undertaken at reduced powers and short circuit loads in order to

accomplish this calibration. Should the Faraday-effect monitor

survive at higher power levels used for operational shots with

plasma loads, it will be incorporated into the apparatus as a regular diagnostic for routine measurements. As discussed above

such a Faraday effect monitor is an absolute instrument whose

calibration depends upon a fundamental property of the rotative

medium. Furthermore, the limiting time resolution of such a monitor

is given by the transit time of light around the closed optical

path.

We initially considered the use of an optical fiber as the

rotative medium as was done in references [2] and [3], but rejected

this approach because of the fiber birefrigence difficulty discussed

above as well as the low value of the Verdet constant of the fiber

core material. Such a low value of Verdet constant would limit the

number of cycles of Faraday rotation and hence the accuracy of the

measurement as discussed above. The use of optical fibers to

transmit light to the Faraday probe from the source and back from

the probe to the detector is, however, very convenient. Such an arrangement provides the necessary electrical insulation of the

probe from ground and eliminates the need for optical alignment

during installation. The light source, detector and recording

equipment could be placed in a screen room at some distance from the

pulse generator, thus, minimizing electrical noise. Since the

monitor is provided with polarizers at the entry and exit of the

measurement light path, the birefringence and residual Faraday

rotation in the fiber due to stray fields play no role in the final

resul t.

We selected as the Faraday rotative medium a high density flint

glass, Schott type SF-58. This glass has a Verdet constant of 0.082

and an index of refraction of 1.907 at the wavelength of the helium

neon laser, 6328 A. The glass has a very low residual birefrigence

and a high transmission for the He-Ne wavelength. An alternative

choice might have been one of the Terbium-doped glasses manufactured

281
by the Hoya Company. These glasses have Verdet constants of the

order of 0.2. We rejected the Hoya glasses both because they are very expensive and because their Verdet constants are temperature

sensitive since the glasses are paramagnetic. While the diamagnetic

glasses such as SF-58 have lower Verdet constants, their values are essentially constant independent of temperature variations. Since

the current required to produce a rotation of 180° is ^10 5 A for

SF-58, the number of cycles of Faraday rotation at the operating

peak currents of 2-4 MA will be 20 or more. Thus, a larger value of

Verdet constant, while useful for reducing the relative error of the

measurement is really not needed.

The closed light path within the rotative medium is realized by

multiple total internal reflections (TIRs) within the rotative

medium. As is well known a TIR introduces a phase shift between the

components of polarized perpendicular and parallel to the plane

incidence. Hence, if the incident light is plane polarized the

reflected light becomes elliptically polarized. This ellipticity in

the incident light would lead to a Faraday signal in which the

amplitude as well as the phase varied with time unlike the simple

expression (Eq. 8). The computer inversion of the Faraday signal

would thus be very complex.

A very simple solution to the problem posed by the ellipticity

in polarization induced by TIR was found by Kard [6]. He showed

that a coating on a TIR interface one quarter wavelength thick

(corrected for angle of incidence) whose index of refraction was the

geometric mean of that of the substrate and the external medium

(usually air or vacuum) would compensate the phase shift between the

parallel and perpendicular components of polarization. Thus, plane

polarized light of arbitrary azimuth incident on a TIR surface

equipped with such a coating would be reflected as plane polarized

light and could be completely extinguished with an analyzer. This

solution is applicable only to cases for which the angle of incidence on the substrate-film interface is less than the critical

angle, i.e., part of the light must enter the coating and be totally

reflected from the coating-vacuum interface. For the SF-58 glass of

refractive index 1.907, magnesium flouride of index 1.38 is almost a perfect match, besides being a very satisfactory material for making

durable optical coatings of precisely controlled thickness. The

critical angle at the coating-substrate interface is arc sin (1.38)"

or 46.4°. Thus, a coating of MgF will effectively compensate the

ellipticity for an angle of incidence of 45°. We have tested such

coatings at 45° incidence and found that a very good null of the

reflected light can be obtained with an analyzer for any arbitrary

azimuth of the incident polarization.

We constructed a Faraday probe in which the light path is along

the sides of a square. The light path was realized by TIR from

three surfaces at a 45° angle of incidence, the reflecting surfaces

being equipped with MgF£ coatings to compensate ellipticity. In

282
order to accurately align the surfaces they were formed by plane- convex lenses which were cemented to a ring of SF-58 glass 6k in.

OD, 3h in. ID, and 5 mm thick.

The light was conducted to and from the ring by optical fibers

equipped with lenses to provide a parallel beam of ^5 mm aperture.

The light was coupled into the ring by a crown glass prism cemented

to a plane convex lens similar to those used for the reflecting

surfaces. Total attenuation introduced by the ring and fibers was

less than 25 db. Figure 2 is a diagram of the device.

COUPLING

Figure 2. Diagram of Faraday effect current monitor.

Figure 3a shows a photograph of the ring in its plastic protective housing with mounts for the optical fibers and polarizers.

Figure 3b is a photograph of the housing with the cover removed to

show details of the ring itself.

283
Figure 3. Photograph of Faraday effect current monitor with

(upper) and without protective housing (lower).

The light source was a 5 MW He-Ne laser. The laser beam was focused into the end of the fiber with a microscope objective. The

detector was a TIED55 avalanche photodiode followed by a transimpedance amplifier with a 10 kfi feedback resistor. The light from

the fiber was rendered parallel by a lens and passed through an interference filter of 60 A bandwidth and then focused by a second

lens on a photodiode. The interference filter served to eliminate

stray light from the spark gaps that switch the test current from a condenser bank.

3. Results — The Problem of Mirror Reversion

The previously described Faraday monitor was tested with the

pulse current supplied the discharge of a 40 kV-37.5 ufd condenser

bank. The peak current of the first half cycle was 326 kA. No

Faraday rotation was detected at all despite a peak current over

three times that required for a full cycle of Faraday rotation based

on published values of the Verdet constant.

284
This decisively negative result lead us to reexamine the entire

concept and led to the realization of a fundamental defect in the

existing monitor which is correctable; the effect of mirror reversion. The easiest way to understand the effect is by the use of the

Jones calculus [7]. This calculus treats the effect of various

polarization components such as rotators or retarders on polarized

light by representing the polarized light by a two-row column vector

which is operated on by 2 x 2 matrices representing the optical

components in the light path. The effect of a number of components

is obtained by multiplication of the series of matrices representing

the optical train. We need only two of these matrices for the

present discussion; a rotator matrix S(a) representing a rotation

of the plane of polarization through an angle a and a retardation

matrix G(y) representing a retardation by a phase angle y of one orthogonal component relative to the other. These matrices can be

written as

a - ia0 sin a

2

(10)

y/2 + i 3

sin y/2

where and are two of the well known Paul i spin matrices.

°c "C -?)

S{a) = cos

G(y) - cos

The effect of mirror reversion can be taken into account by

ascribing the matrix of a half-wave retarder to a mirror; i.e., a mirror is represented by G(tt) . The net effect of a number of

Faraday rotations are interspersed by mirror reflections is given by

S( a] ) G(tt) S(a

2

) G(tt) S(a

3

) G(tt) . . . (12)

Now the following identity follows from the anti-commutive

property of the Pauli Matrices

S(a) G(tt) = G(tt) S(-a) . 285

(13)
Using this identity and noting that G(2tt) is the unit matrix

we have for expression (Eq. 11)

S{a

} ) S(-a

2 ) S(a

3 ) (-^J. . . (14)

Thus, if the total light paths in the Faraday medium be decomposed

into segments between mirror reflections and the segments be num- bered sequentially the sign of the Faraday rotation changes with the

parity of this number. We can thus write for the Faraday rotation

instead of Eq. 1

0 = V /S.dl- ft d£ (15)

odd even segments segments

The reason for our negative result is now obvious. Since the even and odd numbered segments are of equal length in a square path

formed by three reflections the net Faraday rotation is zero.

The solution to the problem of mirror reversion lies in first

making the length of the even numbered segments of the light path as short as possible and second arranging the direction of these even numbered segments as nearly as possible at right angles to the local

field direction so that the integrand of the second integral of

Eq. 13 tends to vanish. Figure 4 shows such a light path and the

glass configuration which will satisfy these requirements. The odd

numbered segments form a closed square in the plane normal to the

current direction and the Faraday rotation due to these should be

just that of Eq. 2. The even numbered segments are then parallel to

the direction of the current so the magnetic field will be normal to

the direction of the light path and hence these segments should make

a negligible contribution to the net Faraday rotation.

If the plane of the ring be tipped at an angle 3 with respect

to the current direction the relative error E, will be

i

£

= (1 - cos 3) + -—^

sin 3 (16)

xodd

As £even

is the order of 5 percent of or less for a typical

construction a substantial tilt can be tolerated without impairing

the accuracy of the measurement.

286
Figure 4. Proposed Faraday effect current monitor modified

to eliminate mirror reversion effects.

The existing Faraday ring could be modified by cementing Amici or roof prisms (45°~90° isoceles prisms) to the reflecting faces

with the roof direction in the plane of the ring. Unfortunately,

the single quarter wave coating of Kard cannot be used to compensate

the ellipticity for now the angle of incidence on the prism is 60°

greater than the critical angle. In principle, the ellipticity can

be compensated for any angle of incidence and any index of refraction

of the substrate by use of Heperin coatings, i.e., symmetrical three

layer coatings of low and high index of refraction [81. The specification of the Heperin coating for a given angle of incidence and

substrate index requires, however, a rather complex numerical calculation.

4. Future Work

A Faraday rotation of the configuration of Figure 4 will be

constructed of prisms of SF-58 glass and either cemented together or clamped in optical contact by a frame of nonconducting material.

The latter procedure would have the advantage that the ring could be

temporarily split for installation around a conductor with inaccessible ends. The large apertures and normal incidence would simply

the use of full-transmission polarizers such as calcite rhombs which

will greatly improve the transmission over that of polarizing films.

This improved transmission may be necessary for very rapid measurements to insure an acceptable signal-to-noise ratio. The large

clear apertures would facilitate the use of mutiple series light

paths to increase the sensitivity of the monitor.

287
An attempt will be undertaken also to compute the required

Heperin coatings for the ellipticity compensation of a roof prism

using the procedure of reference 18]. The existing monitor can then

be made functional by the addition of these prisms to the reflecting

surfaces.

5. Acknowledgements

We wish to acknowledge many valuable discussions with

John Shannon and Norman Rostoker of Maxwell Laboratories, Inc. The

skill and dedication of Leslie Vargady of Metrooptics Co., Glendora

CA, in fabrication of the optical components is greatfully acknowledged. We also acknowledge the devoted services of Lawrence

Houghton and David Lischer of Maxwell for their assistance with the

experimental work and to Clifford Fowler also of Maxwell for developing the algorithm for the numerical inversion of the Faraday

signal. This work was carried out under Contract No. DNA001 -79-C-001

9

of the Defense Nuclear Agency.

6. References

[ 1 ] G.

1_4

A. Massey, D. C. Erickson and R. A. Kadlec,

2712, (1975).

Appl ied Optics

,

[2] A, M. Smith, Applied Optics 17, 52, (1978).

[3] A. Papp and H. Harms, Applied Optics 19, 3729 (1980).

[4] H.

H.

Aulich, W. Beck, N. Douklias, H. Harms, A.

Schneider, Applied Optics 19, 3735, (1980).

Papp and

15] H. Harms and A. Papp, Applied Optics 19, 3741, (1980).

[6] P. G. Kard, Optics and Spectroscopy 6, 339, (1959).

171 R. C. Jones, J. Opt. Soc. Am. 31, 488, (1941).

18] P. Mauer, J. Opt. Soc. Am. 56, 1219, (1966).

288
CAVITY CURRENT MONITORS*

J. Shannon, E. Chu, R. Richardson

M. Wilkinson and C. Trivelpiece

Maxwell Laboratories, Inc.

8835 Balboa Avenue

San Diego CA 92123

In measuring short (^10 ns risetimes), high (^MA) current pulses,

a single turn B probe is useful only when the current path is

precisely known. On the other hand, self-integrating resonance-free

Rogowski coils are relatively difficult to build.

We present a new type of current monitor that is predictable,

simple in construction and immune to capacitive pickups. These

monitors are basically cavities machined in the current carrying

conductors. We will discuss the design criteria and the high frequency

response of these cavity current monitors along with the late time

effects due to resistive losses in the wall of the cavity.

*Work supported by the Defense Nuclear Agency.

1 . Introduction

The development of high power pulse generators has led to the

need for a current probe that is capable of measuring fast (^10 ns risetimes) mega-ampere current pulses. Although these current pulses

are often constrained to flow in current carrying conductors of known

geometries, the actual current distribution in the conductors is not

necessarily known. The cavity current monitors we are reporting

allow one to precisely measure sub-microsecond, mega-ampere current

pulses with high insensi tivi ty to the current distribution. These

monitors are mechanically more rigid, more immune to capacitive

pickup than conventional B probes. Also, they are much easier to

construct than self-integrating, resonance-free Rogowski coils.

Moreover, the sensitivities of these monitors can also be calculated

fairly accurately once the physical dimensions are known.

Basically, the cavity current monitors are re-entrant cavities

machined in the current carrying conductors as shown in Figure 1.

289
For short pulses, the current in the conductor essentially flows as a surface current. As a result, the re-entrant cavity develops a voltage drop across itself corresponding to the rate of change of

flux in the cavity. For longer pulses, the response of the monitor

will change due to the diffusion of magnetic flux into the cavity

wal Is.

Figure 1. Sectional view of cavity current monitor.

In the following sections, we will discuss the theory of operation, the high frequency response, the low frequency response, and

the design considerations of the cavity current monitors.

2. Theory of Operation

As shown in Figure 2, the cavity walls along with the connecting

lead of the 50 ft resistor defines a magnetic pickup loop. An array

of magnetic pickup loops is generally used to sample the flux change

in the cavity at different points around the current path. The

signals generated by these pickup loops are combined using cables of

the same length at a summing circuit as shown in Figure 3. The

combined output is then integrated using a passive RC integration

network to yield a signal proportional to the current.

290
GAP

4-

ARROWS DENOTE CURRENT FLOW ALONG SURFACE

2A

I T~

'CONNECTING

LEAD TO 50

n

RESISTOR

Figure 2. Expanded view of cavity,

CHARACTERISTIC IMPEDANCE lr

CONNECTING CABLE

LOOP

AREA A/

V N LOOPS

SUMMING CIRCUIT

PASSIVE INTEGRATOR

D-t—vw—j—. OUTPUT SIGNAL

SIGNAL CABLE TO SCREEN ROOM v-iTERMINATING

RESISTOR Z„

Figure 3. Equivalent circuit of current sensor.

291
In the circuit diagram shown in Figure 3, it is seen that each

loop is connected to a cable through a 50 ft resistor and the cables

are connected in parallel in the summing circuit. The purpose of the

resistor at the loop is to absorb signals coming back on the cable

from the summing circuit thereby preventing oscillations. Because of

this termination we can separate the connecting cable into two equivalent circuits as shown in Figure 4. In Figure 4a, the wave launched

onto the cable is and is nominally given by

1 A

dB

2

rt dt (1)

where A is the area of the pickup loop and B the magnetic induction

at the loop. At the other end of the cable the remaining N-l

connecting cables and the signal cable constitute a load of Z /N

as shown in Figure 4b. From this equivalent circuit the voltage in

the signal cable, V

2

, as a result of V-j is given by

V

2 = _L_ A

dB

N + 1 dt (2)

(A)

V

o = S

o (CABLE) w = I a e£ v l 2 dt

(B) A2N+1

Figure 4. Equivalent circuits of loop/adder circuit link.

By superposition, the signals from all the loops can be combined

to give on the signal cable

+ 1

N A I

i = 1

d B

i dt

(3)

292
j. u where is the field at the i position.

If all the loops are spaced a distance Ail apart, then Eq. 3 can

be written as

uV = i iL. 5— y R A£ (4)

s N + 1 A£ dt . f t 1

The sum can be approximated by

N r

_ £ B

i Ail j> Bd£ = y

Q

I (5)

when the variation B over distance Ail is small. Eq. 5 can now be

rewritten as

v

- y

o A dl

(ft) s " (N + 1) A£ dt

{b}

The signal cable output is integrated and the voltage out of the

passive integrator is given by

1 u A

V

= RC (N +°1) A£

1 (7)

where RC is the time constant of the integrator,

3. High Frequency Response

For a magnetic probe operated in the differentiati ve mode, the

risetime, tR

, is roughly the self integration time of the probe

0

293
where L p

is the inductance of each loop. The inductance each loop

sees depends on the geometry of the cavity for the particular case shown in Figure 1.

I O § 0 n K_

p

~

0 R 2-u Na

N

where R is inner radius of the outer conductor, a the radius of

the connecting lead to the 50 U resistor (see Figure 2).

Consider an example when R = 0.5 m s a = 0.05 cm , £=0.1 cm, s = 1.0 cm , and N = 6 . The probe inductance is

approximately 126 pH giving a risetime of 2.5 ps.

As illustrated in the above example, the high frequency response

of the cavity current monitor for fast, mega-ampere current pulses

should not, in general, be a limitation in its applications.

4. Low Frequency Response

As mentioned previously, the response of the cavity current

monitor to longer current pulses will deviate from that of the high

frequency response. To calculate the low frequency response, consider

the model shown in Figure 5. In the cavity walls, which are assumed

to have a finite conductivity, the electric and magnetic fields

satisfy:

9EW _ 9B

9x at

35T " ~V Ew

(8)

where a is the conductivity and x is the distance into the wall.

In writing these equations, we have made the problem one dimensional.

This is a good assumption if the thickness of the current sheath 6 is small compared to the cavity dimensions. This thickness 6 is

roughly

which is the distance the field can diffuse in a time t 294
For aluminum

and if

then

- = 4 x 10" 6

Q - cm , a

t = 0.1 ys , -? 6^5x10 mm .

Hence, this assumption can easily be satisfied.

Using Laplace transforms B and Ew

are given by

-ax

B = y

Q

I Q

(s) l

a I (s)

r _ o

v

' -ax

Ew a

e

(9)

where

a =

/y Qas

and I Q

(s) is the current flowing around the cavity per unit width

To find the response of the monitor we use Faraday's law

3t^

integrating around the boundary (x = o) of the cavity. At the walls

we pick up a contribution from E w

(o) . The gap voltage is (ignoring

signs)

I a

v = -°_ D + \i n

l n As

g o

(10)

J o 0

D being the distance around the loop and A being the area. We can write this as

295
V

g

= %

! Q

As 1 +i r

1 (ii)

TOTAL LENGTH AROUND LOOP

AREA A

Ew

= ELECTRIC FIELD IN THE WALL

J = CURRENT DENSITY IN THE WALL

B = MAGNETIC INDUCTION

V

6

= VOLTAGE ACROSS THE GAP

Figure 5. Model for low frequency calculations.

Let us now assume the current is a unit step function (I 0

= 1/s)

and that we have a perfect integrator (multiply by 1/s) and let us normalize to the high frequency response (divide by y0 A). We then

get for a normalized output signal (from the integrator)

V(s) =i 1 + (12)

In the time domain, this becomes

4

? °

2

1 (13)

or

v ( t) - 1 (y

296

(14)
where

Here, t

D plays the role of an effective cavity diffusion time.

\, The term (t/t

D

) 2 in Eq. 14 represents an error term in that it

is a deviation from an ideal step function. The error can be quite

significant for late times. In passing, it can be noted that t~ is

a minimum for fixed A if the cavity is a square.

Since the error increase with time, the droop caused by the

finite time constant of the integrator tends to compensate. The

effect of the finite integration time constant of integrator can be

determined by replacing 1/s with l/(s + 1/RC) in the front of

Eq. 12. This gives

where RC is the time constant of the integrator. In the time

domain,

5. Wall Resistivity Compensation

(15)

V(t) = I

t_

RC

1 +

A

2 (16)

for t < RC , the integral can be expressed as

V(t) = I

t_

RC

1 + (17)

297
from Eq. 17, we see that the exponential term tends to compensate for

the diffusion term. For an aluminum cavity current monitor with

A = 0.1 cm2 and D = 2 cm , is approximately 8.7 us. The

percentage deviation from a step response for various values of RC

is shown in Figure 6.

Figure 6. Deviation from step response of rectangular beam

Faraday cup for various integrator time constraints.

6. Conclusion

We have described the characteristics of a current probe that

is capable of measuring mega-ampere current pulses with sub-nanosecond

risetimes. The construction of the cavity current monitor is simple.

Although the response of the probe changes with low frequency signals,

the effect is well understood and can be minimized by using materials

with high electrical conductivities along with a square cavity. This

effect can be further minimized by choosing integrators with appropriate RC time constants for each application. Moreover, this

effect can be totally corrected using digital computer techniques [1].

298
7. References

R. Richardson, et. al . , "Calibration of the BLACKJACK 5 Pulse

Generator Output Power", this conference.
Discussion Session 5

Richard MaTewski (IREQ). To clarify my question which ended the

previous Discussion session, I think that it is more practical to

make a drawing on the blackboard (see figure 1). The question was to the design of an impulse current transformer such as manufactured

by the Pearson Company. If that stands for a section of the winding

of the transformer, we can distinguish some elements in it. There

is a line of resistors which are in the kilo ohm range. There are taps

of the winding connected to the resistive wire which probably constitutes

the distributed burden and there are extensions of this, legs separated

by a thin milar foil from a metal copper band and this increases

capacitance between these legs and the ground foil. On top of this,

the whole winding was terminated by a fairly complex resistor network

which I should admit was difficult for me to understand why. So, if

this question can be answered, I would appreciate it. It would cast

some light on the particularities of behavior of the Pearson transformer.

There is perhaps another element in the case of coils made

without magnetic core. I feel that one can improve the bandpass of

the coil by using tape rather than wire to produce the winding. The

idea is to reduce the characteristic impedance of a line formed by

the helix of the winding and the external metal shield. Such lines

can be matched by an appropriate resistor at the end if it is kept

within a few hundred ohms. I don't know whether this idea is viable

and to what other comments people manufacturing these coils might

make.

Marco Pi Capua (Physics International). Well, our experiences have

been with rather simple devices like the transmission lines that I described. We have not made any attempt either to introduce damping

or to build a complicated structure. In general, Physics International's

design philosophy has been to keep the device as simple as possible

so it can be easily modeled. The transformer under discussion could

perhaps be described as a parallel connection of two transmission

lines. (One is associated with the shield and the other is associated

with the damping network.) The transformer could be modeled with

one of the finite element network analysis codes.

Jon Barth (Barth Electronics). I think that you will find that

something like this would be more amenable to a delay line type of

analysis. The analysis is not quite as simple as has been implied.

It is much more complex when you have the other conductors right next

to each other.

John Shannon (Maxwell Laboratories). OK. Can we leave Rogowski

coils and go on to some of the other topics? Are there any more questions on the papers in the morning?

300
Figure 1. Schematic representation of flux in a current

transformer with off-axis excitation and distributed

load for i 2.

301
Steve Levy (U. S. Army Electronics & Technology Laboratory).

I would like to ask a question of the Raytheon paper on the data

acquisition system. I noticed in the data acquisition system that

they used a current transformer as a method of picking up the signals

which means that they were not dc coupled into their modulator.

When you have multiple switches that are operating parallel, how do

you detect the continuous conduction mode or that mode where the

tube fails to recover without using a voltage divider or some sort

of dc current sensors? That is my question.

Bob Kolibas (Raytheon). It's a good question and the answer is you

don't unless you do have a dc coupled method available.

Levy . The answer I guess is that they do not detect continuous

conduction. We have a paper we will be reporting on at the pulse power

conference in Albuquerque where we do treat the problem of continuous

conduction. So, I will wait until that meeting to discuss this topic.

Pi Capua . I have one question about the monitoring of current

densiti es with a scintillator. Does scintillator material dendrite

at all when you deposit a large number of electrons in it?

Bob Scarlett (Los Alamos). It does after a number of shots. At

these current densities (50-100 mA/cm^ for 5 ys), we can get twenty

or thirty shots before it dendrites or starts to turn yellow due to

radiation damage. We have also noticed that there is an electret

effect in the plastic.

Pi Capua . What provisions do you make to remove the large amounts of

charge from the back of it? Do you have a collector behind it?

Scarlett . I think that most of the charge is removed by the plasma

itself. The beam plasma in the gas tends to discharge the surface

charge. The deep charge doesn't come out except from a slow diffusion.

Pi Capua . How many joules per square centimeter are incident on the

film that you are using?

Scarlett . There is about 50 milliamps per square centimeter and the

pulse length that was actively contributing the light was probably

on the order of a microsecond or so and these are 350 kev electrons.

Shannon . I think Uglam and Cooperstein did this for total current

monitors in about 1970 or 1972 and one of the comments that was made

to me was that in order to get away from the scintillation problem,

you had to use vacuum UV grade fused silica. All plastics were claimed to have scintillation light that was not Cerenkov light.

Would you make a comment on that?

302
Scarlett . I used to do research which involved the detection of

relativistic heavy ions. One of the things that we did at that time

was to look at the scintillation properties of various glasses and

plastics using relativistic heavy ions at the Bevalac. We determined

that the scintillation properties of plexiglass were very low, not more than a percent or so of the Cerenkov signal.

Neville Harris (Ion Physics). I would like to ask the reproducibility

of the lucite. For example, was the Cerenkov radiation dependent on samples, impurities, or degree of polymerization? I would like to

compare your method with the use of blue cellophane.

Scarlett . The Cerenkov effect is a classical, collective optical

effect . It doesn't depend on any of the microstructure or the

molecular physics of the material. It depends purely on the index of

refraction of the material. Measurements that I made when I was

doing particle detection work indicated that it usually was very

consistent.

Jack Katzenstein (Maxwell Laboratories). I have a question for

Dr. McDaniel about his very interesting piezoelectric pressure probes

for measuring current. If the probe is placed too close to the plasma

source, isn't there going to be an appreciable pressure contribution

from the ablation from soft x-rays?

Dfl lion McDaniel (Sandia). The answer to that is yes there is. The

way we run our plasma implosion source, the plasma does not implode

until long after current peak. So, we use the probe to verify the

total current delivered to the load and there is no problem because

of the separation time between current peak and radiation from the

plasma in our experience. That may be different for other people who

implode on peak current though.

Katzenstein . Well, the only comment I would make to that though is

how do you optimize coupling or is this a problem with you?

McDaniel . For a low inductance diode which is like inductances of

/ nanohenry or less, imploding after peak current is the optimal for

coupling kinetic energy. Other labs may not have seen this since no other labs have accelerators with as low inductance diodes. So I will be glad to discuss that with you after the meeting. Since this

meeting is not on plasma implosion, I think I would not want to take

the time here.

Shannon . This is a sensitive subject and let's not get too far into

this implosion business. We will all get into trouble. Are there

any other questions?

Ed Chu (Maxwell Laboratories). Yes, I guess I would like to ask a question to Dill ion. You mentioned that the pressure transducers are very good for high magnetic pressure. Can you comment on the other

303
end of the application? At how low a current should I stop using

that probe and how long a pulse would be really convenient for the

application of these transducers?

McPaniel . It depends on the situation. If you are in a strip

transmission line where the B is uniform, then you can get more signal

out of the probe by going to a larger diameter area since the probe

output goes up as area. I feel that you can probably use this type

of probe down to where you have magnetic pressures on the order of 50

or 100 bars provided you can make a large enough area probe that you

are over your signal-to-noise ratio from EMP. So, you have to look

at each specific case. The paper will have piezoelectric crystal

constants in it for quartz. There are other piezoelectric materials

that have higher sensitivities and we are investigating lithium

niobate at the moment although for some reason it doesn't seem to

work right.

Pi Capua . I would like to make a comment about that. In 1978, we performed a magnetic insulation experiment at PI (I = 1 MA,

r = 6 x 10~2m, B = 3.3 T, p = 44 bar) and Vern Bailey, a colleague at

PI, suggested I use quartz gauges for current measurements. I turned

down the suggestion because of the low signals we would obtain. This

is the reason we used the shunt approach instead.

John Shipman (Sachs Freeman/NRL). I would like to ask Dillion what

the indications were that the B probe holes were flashing over?

McDaniel . If you look closely at the probe holes, you can see "crow

feet" tracks in the radial direction of the outside of the hole to the

inside of the hole. Of course, we also lost signal. The B signal

would go up, come down and all of a sudden would drop to zero which

indicates that something is flashing. Inspection of the hole showed

"crow feet" tracks that started on the outer radius of the hole going

in towards the radial direction and simultaneously there would be a burn mark within a centimeter radius of where that hole was located.

But, I can't explain why it happened because I can't really calculate

the voltage drop across the hole between the quarter-inch diameter

sitting out at a radius of five centimeters even though there are those enormous dl/dt's.

Pi Capua . I would like to share some data on the effect of penetrations

for diagnostics on magnetically-insulated vacuum transmission lines.

After a few pulses (20-30) we noticed a very severe erosion, in the

vicinity of the penetration, downstream from it. This erosion, which

gave the penetration the appearance of a keyhole is shown schematically

in figure 2 below and photographs appear in figures 3 and 4. The

erosion is probably caused by perturbation of the space charge electron

flow due to the absence of return currents in the vicinity of the

hole. The figures contain all the experimental details.

304
(a) Faraday Cup

( + ) 4 0.5 cm

material

removed

< + )

Power Flow

(-) Boundary

Current

(-)

Figure 2. Schematic of anode penetration for magnetically

insulated transmission line diagnostics (a) side view

(b) view from cathode.

305
2 cm

POWER

FLOW

Figure 3. Anode erosion caused by penetration in anode wall.

View is from cathode, inside transmission line. Hole was circular at first. Erosion was caused by 30 shots.

(M. DiCapua and T. S. Sullivan, Proc. 2nd IEEE International

Pulsed Power Conference, Lubbock, TX, IEEE 79CH 1505-7, 1979).

306
2 cm

POWER

FLOW

Figure 4. Anode erosion caused by penetration in anode wall.

View is from outside the transmission line. Hole was circular at first. Erosion was caused by 30 shots.

(M. DiCapua and T. S. Sullivan, Proc. 2nd IEEE International

Pulsed Power Conference, Lubbock, TX, IEEE 79CH 1505-7, 1979).

307
Harris . Was that arrow you drew in the direction of the magnetic

field? The direction of the magnetic field rules out a multipactor

mechanism.

Bob Hebner (NBS). I would like to know how soon Maxwell anticipates

having a magneto-optic current monitoring device operational?

Katzenstein. We have ordered the glass, and if the optician doesn't

think that it is too difficult a job to do it, we ought to be testing

it within three to four months.

Malews'ki . We have been playing with magneto-optic current transformers

for some time. And there were finally two elements which decided

that we abandon these particular techniques. Perhaps it is wise to

show where the particular problem was found. The idea was to monitor

current in high voltage transmission lines at high potential and what we did was to have a slab of lead glass placed in the vicinity of the

current carrying conductor in such a way that the magnetic field of

the current was along the axis of the glass slab. A polarized light

beam was transmitted through the slab and deflected down to ground

potential to monitor the current as expressed as the rotation of the

polarization plane of polarized light. The glass slab was approximately

10 inches long and roughly 1/4-inch diameter. Unfortunately the heat

generated by the current circulating in the conductor produced some

local temperature gradients in the glass probe in turn producing

mechanical stress. The consequence was that the piece of glass became

bi refringent . The linearly polarized light beam which came out of

the slab was not only twisted by a certain angle but was also

elliptically polarized due to the residual birefringence induced by

thermal shock. Recently, I heard some encouraging news. People have

abandoned the idea of using massive glass slabs which are naturally

prone to this effect, and are thinking in terms of using optical

fiber wound around the current carrying conductor. The beauty of

this solution is evident. By increasing the number of turns, one can increase the sensitivity of such a modulator because the twist of the

polarization plane due to the measured current would be proportional

to the path of the light passing in the several turns of this coil

and, therefore, several revolutions of the polarization plane can be

achieved even with a moderate current. The problem here is that when

one starts to twist fiber optics, birefringence will be produced by

the act of bending. This apparently can be compensated by twisting

the fiber in the process of winding the coil and it has been reported

that such coils were built which are free of birefringence. If it is

all true, this type of coil fixed in some sort of epoxy block could

be an excellent magneto-optic modulator. Even if there was some residual thermally induced birefringence it will be much smaller than

before for two reasons. First of all, fiber is thin, therefore the

stress due to the thermal effects would be smaller and, secondly, the

signal, due to the number of revolutions of the twisted polarization

plane would be much stronger in comparison to the effect of birefringence.

308
Katzenstein . First I would like to thank Dr. Malewski for bringing

out or reiterating the points that I made in the first few minutes of

my paper. We have looked at this. One of my co-authors, Bill Caton,

is a fiber optics expert and it is his opinion that a polarization

preserving fiber is beyond the present state-of-the-art. Now, this

can work, if you are prepared to make measurements of very small

polarization rotation angles based on differential intensity

measurements. I think you made a bad choice of glass, incidentally,

because there are the SF58 and certain other Schott glasses which

have an extremely low strain birefringence. Furthermore, they also

have extremely low loss tangents. I think also that another disadvantage

of this scheme is that it does not use a closed light path which

means it is prone to changes of position or orientation of the rotating

medium with respect to the conductor. If you have something dangling

from a transmission line, this is apt to happen.

309
Session D - Data Acquisition

HIGH-VOLTAGE MONITORING AND CONTROL THROUGH FIBER OPTICS

John G. Stewart, Jr.

William A. Petty

Harry Diamond Laboratories

Abstract

Recent advances in the field of fiber optics have enhanced our ability to monitor and control high voltage in a hostile electromagnetic

pulse (EMP) environment. Harry Diamond Laboratories has utilized these

advances to pursue the development of a Remote Command Data Link (RCDL)

for the Army's threat-level EMP simulator. This simulator utilizes twin

Marx generators which can produce output voltages in excess of +3 MV.

Through the use of the RCDL, this simulator can be remotely fired and

real-time oscillograph records made that represent the +3 MV voltage

waveforms that are impressed across the simulator peaking capacitors.

The RCDL was designed to allow normal pulser diagnostic maintenance,

Marx output waveform monitoring, and adjustment of the Marx timing

signals from a command and control center located approximately 30 m from the pulser. Specifically, the RCDL features synchronized command

firing of the bilateral Marx generators, performance check of the pulser

high-voltage trigger system, Marx generator erection waveform, advance

or delay of a single Marx trigger for optimum simulator performance, and

interlock with the simulator console for the arm and abort function.

The RCDL has recently been made an integral part of the Army's

threat-level EMP Simulation Facility, located at Woodbridge, VA. This

system is presently being evaluated for long-term-performance.

Preliminary data reveal that the RCDL provides the desired design

features with reliable performance.

This paper presents an overview of the RCDL with emphasis placed on its usage in a particular EMP test facility. A follow-up paper by the

Physics International Company presents a detailed circuit design of this

system.

310
1. Introduction

Recent advances in the field of fiber optics have enhanced our ability to monitor and control high voltage in a hostile electromagnetic

pulse (EMP) environment. Harry Diamond Laboratories has utilized these

advances to pursue the development of a Remote Command Data Link (RCDL)

for the Army's threat-level EMP simulator. Before discussing in detail

the unique characteristics of the RCDL, the need for developing this

type of instrumentation will be established. This introduction will

focus on the historical development that led to this effort, with an explanation of how the RCDL relates to the high-voltage system that it

supports. Some general background information will be given on the

complex high-voltage electronics used by the specific EMP simulator that

interfaces with the RCDL.

The need for EMP simulators and, hence, the RCDL can be understood

by focusing on the electromagnetic environment that is generated when a nuclear device is detonated in the exoatmosphere. This detonation

generates an EMP which propagates over a vast geographical area. The

nature of this pulse makes it potentially destructive to many electronic

systems, including a significant number of defense related systems. The

present restriction on testing of nuclear devices in the atmosphere

created an urgent need for simulating this electromagnetic environment

in order to empirically investigate the susceptibility of certain

military electronic hardware.

2. Approach

The Army has filled the need for generating a simulated EMP

environment through the development of an EMP simulator that hereafter

is referred to as AESOP, Army EMP Simulator Operations. This facility

uses state-of-the-art high-voltage components, electronics

instrumentation, and calibration techniques to achieve the high field

level experienced by the real EMP threat. Because of the very high

voltages developed in this simulator, reliable and efficient monitor and

control methods for the AESOP pulser are constantly being pursued. For

example, the AESOP pulser can generate a voltage in excess of 6.5 MV

accompanied by an energy of over 60 kJ across a single high-voltage

output switch. With voltages and energy of such large magnitudes, a critical requirement is timing synchronization between command pulses.

Two generators are required to erect to their full potential, and the

energy is delivered to a single output switch within a time window of

approximately 70 ns. This event must be accompanied by a trigger system

timing jitter of ±10 ns. The ±3.5 MV impressed across the pulser

output switch determines the radiated output of the simulator antenna.

It is the need to measure and monitor each polarity of this voltage

in terms of erection time amplitude and timing jitter that prompted the

development of the RCDL.

311
In determining the specifications to be established for the RCDL,

the pulser's operational characteristics were examined closely. Two areas that are perhaps the most critical to the simulator's operation

are the synchronization of high-voltage triggers and individual Marx

generator erection characteristics. Factors that influence synchronization are Marx erection times, Marx erection jitter, and level of

voltage delivered by each Marx generator. Marx erection, which proceeds

very rapidly, is the process by which the Marx generator's spark gaps

are overvolted, usually sequentially, and broken down. Marx erection

time, the time for all Marx switch gaps to close so that load current

begins to flow, is on the order of 150 ns for the 35-stage Marx

generators in the AESOP. During erection, the Marx generators pulse

charge their respective peaking capacitors in about 65 ns to the

required high voltage (1 to 3.5 MV of opposite polarity for each

generator). After the generators erect, the output switch breaks down,

discharging the peaking capacitors and Marx generators into the

antenna. Fig. 1 shows actual Marx generator waveforms and illustrates

the electrical parameters of the bilateral Marx generator configuration.

Trigger synchronization is another important feature considered when

establishing design characteristics for the RCDL. To insure that the

two Marx generators will erect simultaneously, the individual Marx

generator stages for each side of the pulser must be triggered with a very low timing jitter. During this erection time, the pulser peaking

capacitors are charged to the peak voltage of the Marx output. This

voltage is simultaneously delivered to the pulser output switch. This

output switch is self-breaking and is designed to close at 90% to 100%

of the peak output voltage from the individual Marx generators. With

this closure, the peaking capacitors supply the initial fast-rising

pulsed voltage across the closed output switch to the antenna. This

voltage can be used to analyze the performance of the pulser. However,

to acquire knowledge of this voltage waveform and output amplitude, we must have the capability to sample voltages in excess of 3 MV with rise

times in the nanosecond range. Thus, we have established a real need

for the subject RCDL.

Through the use of the RCDL, it was anticipated that one could

monitor the real-time critical operation characteristics of the AESOP

simulator. This information on the pulser output voltage could help

troubleshoot pulser operational problems. The voltage waveforms could

reveal the operational characteristics of the individual Marx

generators. This information would provide insight into any problems

that may have occurred during pulser firing, as well as establishing a record of the pulser's normal operational performance normality.

3. Significance of RCDL Data on Operations and Maintenance

Because of the critical timing and synchronized erection requirement

for the Marx generators, diagnostic maintenance is required for any

312
Side 2

iiiiiiiiii

Side 1

bmbbbmbbm

Ls Rs

Cg

Output Switch

Cp -1- S Rsh Cg

Cg = Marx generator

Ls = Stray series inductance

Rsh = Shunt resistance formed

by the Marx generator

spark gap triggering

resistors •

Cp = Peaking capacitor

Rs = Sum of stray resistance

within Marxes

Za = Driven impedance associated

with antenna

Figure 1. Bilateral Marx pulser simplified schematic and

Marx generator output waveforms

313
change in output level or replacement of critical components. The RCDL

has helped save time and simplified diagnostic maintenance and analysis

of the Marx waveform erection times. The Marx waveforms are recorded as oscillographic data output from the RCDL fiber-optic down-link system

each time the EMP is launched. This visual aid, along with the delay

control mode located in the command and control center, adjusts the

erection times of the Marx generators by controlling the trigger timing

synchronization. Nonsynchronized operation can lead to voltage breakdown across one set of pulser peaking capacitors if allowed to exceed

certain bounds. The RCDL down link checks the performance of the

synchronized trigger system. The initial check or coincidence mode

insures that the total trigger system is operational prior to actually

firing the antenna and also checks the simultaneity of the Marx

triggers. This mode is interlocked with the Marx dc charging, and

unless the trigger pulse timing is acceptable within a +10 ns tolerance,

dc charging of the Marxes cannot take place.

The up link or command trigger initiation is located in the command

and control tower. These command pulses are activated through a variable advance or delay control to insure that the proper synchronization requirement exists between the command pulses. The modular

construction and the light-emitting diode (LED) monitor indications of

several events in the timing system have enhanced the troubleshooting

techniques and decreased out-of-servi ce time.

The coarse or initial synchronization takes place by monitoring the

high-voltage triggers. The monitor points, one for each pulser trigger

side, measure the instantaneous current representing the 80-kV highvoltage trigger required to erect the Marx generators. This monitoring

is done with a dual input oscilloscope for synchronizing the two trigger

pulses. Once the trigger pulses have been physically synchronized

within the +10 ns specification by cable selection, onboard or Marx

waveform synchronization is done as described previously.

The technique for measuring these waveforms required time-consuming

tests before the integration of the RCDL to the system. Shield boxes

and data acquisition hardware were placed on board each side of the

pulser. Here, the real-time waveforms could be monitored by onboard

technicians. By observing the waveform erection and breakdown criteria,

the high-voltage triggers could be synchronized through adjusting the

trigger signal delay to one generator relative to the other. This

technique is considered the fine tuning synchronization method.

The RCDL eliminates onboard hardware and personnel needed for Marx

waveform monitoring. The monitoring is done by using electronic

hardware and fiber-optic data links to observe these trigger pulses and

Marx erection waveforms. By built-in adjustable delay lines, the Marx

generator synchronization or fine tuning is simplified.

314
The calibration and the accuracy of these measurements are governed

by Army calibration standards for oscilloscopes and the initial verification of vendor specifications of the fiber-optic systems. To insure

amplitude verification, an external pulse is injected into the fiberoptic data link, and a bias control adjustment is made on the fiberoptic receivers.

4. Reliability

No single design objective, except for the actual simulated pulse

itself, is as important as the reliability of the pul ser/antenna

simulator facility. Current EMP vulnerability data-acquisition methods

require the production of a large number of identical EMP 1 s . Deviations

from this concept extend data processing and interpretation. System

reliability is enhanced by the initial pulser design and development

emphasizing reliable high-voltage techniques.

Prior to interfacing the RCDL with the AESOP pulser, consideration

was given to the hostile EMP enviroment as radiated from the AESOP

antenna. Sound shielding practices were applied to the RCDL commandmodule interconnecting cable that is interfaced with the console used in

the command and control function for the AESOP simulator. Grounds,

where necessary, were bonded to form common grounds. Onboard and RCDL

console modules were contained in shielded enclosures with interface

connections filtered for protection from the electromagnetic

envi ronment.

315
REMOTE COMMAND DATA

SIMULATOR PERFORMANCE

LINK PROVIDES ENHANCED

IN HIGH EMP ENVIRONMENT

Stan Lyons

Physics International Company

2700 Merced Street

San Leandro, California 94577

The Remote Command Data Link (RCDL) system was designed

to provide an advanced method of determining EMP simulator

performance on a shot-to-shot basis.

1. Overview of the RCDL Sytem

The purpose of the RCDL is to command trigger each half

of the AESOP EMP simulator with a low jitter (< 1 ns) pulse

and with an adjustable delay of 0 to ± 25 ns between the two

halves. A signal generated by the output of the onboard

trigger systems is then transmitted to the RCDL, and the

timing is compared for coincidence (± 5 ns). Delay compensation is necessary to provide time-equalized pulses regardless of the delay time setting. The pulses are compared for

conincidence . If they are within the + 5 ns window, an

interlock is activated. This allows dc charging of the

AESOP Marx generators in preparation for the discharge of an

EMP pulse. The self-check of the triggers is necessary to

assure proper trigger operation because of the twin Marx

generator configuration of AESOP. The Marx generators must

be triggered almost simultaneously to avoid overvolting

various components of the simulator. Since the Marx erection time varies slightly with charge voltage and polarity,

an adjustable delay is provided to correct for this variation. The delay adjustment is determined by observing the

Marx generator erection waveforms monitored by E-field

plates on the simulator. These waveforms are then transmitted to the RCDL and to the operator. Overall performance

of the simulator is evaluated from observation of the

waveforms, and any timing corrections are made.

2. Design Approaches - Optical Link

The above conditions required several novel design

approaches. The high EMP environment of the AESOP simulator

and the necessity of total electrical isolation limited

methods of data transmission to optical means. The requirement of transmitting wide bandwidth analog signals and digital timing pulses with low jitter introduced several difficult design problems.

316
The main problem was the transmission of Marx erection

waveforms. If a totally digital system were used, analogto-digital conversion would be necessary. But this raised

the problem of converting wide bandwidth video signals with

good resolution. On the other hand, the one advantage of

digital transmission is that data can be stored on board and

retrieved at the operator's convenience. However, waveform

fidelity was believed to be more important than absolute

voltage value. Therefore, analog real-time data transmission was selected. To accommodate the digital timing

pulses, a video bandwidth in excess of that needed for the

Marx output waveforms was required. For low jitter, a risetime of 10 ns or less was preferred, implying a bandwidth an order of magnitude greater than required for the

analog signal (- 300 MHz versus - 30 MHz). At the time the

RCDL system was designed, an analog optical link with this

capability was not available. It was determined that by

operating the optical link in a saturated mode for the

digital signals, a marked increase in risetime would be

possible. We discussed these issues with Meret, Inc. Meret

believed that it could supply a system to meet all of our

requirements, including an analog bandwidth of 1 MHz to

130 MHz, an operating frequency range of 20 kHz to 200 MHz,

and a system dynamic range of at least 40 dB.

3. Command Trigger Synchronization

The command trigger console system consists of a fastrise pulse generator, a variable-delay generator with tapped

delay lines, two optical uplinks, and support circuitry such

as power supplies. The optical trigger signal is converted

to an electrical signal on the simulator, and this triggers

the AESOP Trigger System (ATRIS). A high-voltage resistive

monitor on the output of each trigger generator supplies a signal to the RCDL onboard optical transmitter. This signal

is shaped to provide a fast-rise, narrow-width pulse (< 5 ns

rise, 20 ns width), which drives the transmitter in a saturated mode. The narrow pulse is necessary because the Marx

generator waveform follows this pulse by about 50 ns. When

the pulse is received by the RCDL console optical receiver,

it is fed through a high-pass filter which blocks the Marx

erection waveform. The pulse is then applied to a highspeed comparator, which is set to a threshold above the

normal wideband noise generated by the optical system. The

comparator provides a logic-level output pulse, which is fed

to the delay generator to remove any delay that was dialed

in initially.

At this point, the two pulses are checked for coincidence through a logic gate. If they are within ± 5 ns, an

output from the gate activates further logic and pulse

stretcher circuitry. This culminates in a side-one/side-two

317
timing O.K

light and an interlock signal to the TEMPS command and control, which allow the Marx generator charging

sequence to start. An additional feature is a misfire

detector activated any time trigger pulses are received

without a command trigger initiation. This misfire detector

automatically aborts any charging or firing sequences of the

AESOP pulser.

4. Marx Generator Waveform Transmission

The Marx generator waveforms are detected by E-field

plates inside the main vessel. These plates are approximately 1 ft^ in area and have an attenuation factor of

approximately 20,000:1, yielding about 150 volts peak into

50 ohms at the normal operating level of AESOP. This voltage is attenuated further to about 1.5 volts (100:1) and

applied to a power tee into the onboard optical downlink

transmitter. Both signals are then transmitted to the RCDL

optical receivers. The receiver output directly drives an

oscilloscope output, and the oscilloscopes record the

waveforms in real time.

As anticipated, several problems were encountered with

the data acquisition and recording scheme which were difficult to solve in a high EMP environment. The output level

of the Marx generator waveforms is normally about 500 millivolts peak, and any leakage of EMP disturbs these waveforms

considerably.

5. EMP Shielding

The AESOP simulator successfully generates a very high

EMP field environment. Because of this, double shielding of

all circuits and optical isolation techniques are used whenever possible. The RCDL console has RFI shielding enclosures for all power supplies, optical receivers and transmitters, and logic circuitry. Optoisolators are used to

interface with the AESOP control console, and internal ac

power wiring and relay logic is kept to a minimum.

Interface with the oscilloscopes is through double-shielded

cable to the oscilloscope screenbox. It is this run that

appears to pick up most of the noise affecting waveform

fidelity. This area can be improved by alternate shielding

methods, or relocation of the RCDL into the screenbox.

Inside the Faraday cage of the pulser structure, the RCDL

onboard electronics are in a much less severe environment.

Even so, the same extensive methods of shielding are used.

6. Operation of the RCDL

The RCDL system streamlines the tedious task of determining whether the AESOP simulator is operating properly.

318
It also makes possible shot-to-shot timing adjustments that

keep the simulator at peak efficiency.

A normal discharge cycle begins by arming the AESOP

simulator from the control console. Five seconds after

turning the arm key, the Fire Ready light on the RCDL console lights up, and the operator presses the Fire button.

This directs the ATRIS to check out synchronization. If the

coincidence criterion is met, the S1/S2 Timing O.K. and the

Armed indicators on the AESOP console light up. Marx charging can begin shortly after the Charge Complete indicator

lights, and the RCDL Fire Ready light will come on again.

Pressing the Fire button this time will erect the Marx generators and launch the EMP pulse. The Marx generator waveforms are recorded as the simulator fires. The operator can

discern from the oscillographs proper synchronization and

overall pulser performance.

7. Future Uses of RCDL Concept

The most exciting prospect for the use of the RCDL

concept is multiple synchronous or ripple EMP wave generation by several simulators controlled by an RCDL system.

The diagnostic problems associated with these simulator

systems would be greatly simplified. The RCDL can monitor

on each shot and eliminate downtime used by traditional

diagnostic methods.

The RCDL could also be employed to control an entire

pulsed power system through fiberoptics coupled to a microprocessor-based data acquisition and control system.

Simulator performance, test subject response, and other

parameters could then be integrated into one set of data for

study and recording. The RCDL could streamline many facets

of pulsed power system testing and operation, while keeping

sensitive data acquisition and measurement systems electrically isolated from high EMP or RFI environments.

Acknowledgements

The author would like to acknowledge the assistance of

J. Stewart and W. Petty, EMP Simulation Group, Lab 1000,

Harry Diamond Laboratories, Adelphi, Maryland; J. Keating

and D. Medved of Meret, Inc., Santa Monica, California; and

J. Dempsey of Physics International Company.

319
A FIBER OPTIC MONITORING SYSTEM FOR ANTARES PULSE POWER SYSTEM*

M. E. Thuot and W. R. Scarlett

University of California

Los Alamos Scientific Laboratory

P.O. Box 1663

Los Alamos, NM 87545

Antares is a 40 kJ CO2 laser system being built for inertial

confinement fusion studies. The electron beam controlled discharge

in the laser power amplifiers is driven by eight 1.2 MV, 300 kJ,

2 microsecond pulse generators. Each electron gun produces a 5 microsecond pulsed beam of 550 keV electrons with 50 ma/cm^ current

density and total beam area of 9.3 m^.

An electro-optic interface to a computer based control and data

acquisition system has been developed to provide a capability for the

measurement and operational monitoring of pulsed analog waveforms and

high voltage system timing information.

Each interface channel consists of a signal powered transmitter,

a fiber optic cable, and an optical receiver which interfaces through

a CAMAC compatible module to the control system. The data channel is

capable of rejecting 600 kV of common mode voltage, and operating in an electromagnetic interference environment of 500 kV/m and 50 kA/m

at 140 kHz.

The analog transmitters send 10 MHz bandwidth amplitude modulated

information to transient waveform digitizers in the data system.

During operation about 100 data channels monitor voltage and current

in each pulser and anode current density in the laser power amplifier.

These devices have also been used to measure electron beam current

density during electron gun testing.

Timing information of 2 ns resolution originates at each stage

of trigger amplification and main pulse output in the ten power

amplifier pulsers. The 120 channels of information are processed to

provide a complete prefire diagnostic of each pulser.

The design, calibration, and performance of this system and its

components will be presented.

Work performed under the auspices of the U. S. Department of Energy.

320
Discussion Session 6

Bob Hebner (NBS). I would like to ask either or both of the first

two speakers about the pulse voltage measurement system that they

have described. They showed what they characterized as an electric

field pick-up, a relatively sophisticated transmission system to get

the waveform back down to ground potential and then reconstruct it.

But, what I was curious about was the design accuracy of that system,

the achieved accuracy, and the method of verification.

Stan Lyons (Physics International). We felt that the waveform fidelity

was much more important than the absolute voltage value. Since this

is a real time system it was easy to compare the reconstructed waveform

on the ground with the on-board diagnostic pictures we were taking

with the screen box. So, we felt that linearity within one or two percent was really not necessary. We were after relative value

changes in waveform from shot to shot as long as they reproduce

within 5 or 10 percent we found that that was quite acceptable.

Bill Boyer (Sandia). I have a question for Stan also. I was wondering

about the quality of the fiber optic data link. Is it a 130 megahertz

analog link, how was it to work with, how did it perform?

Lyons^. The entire system was a system supplied to us by Meret, Inc.,

of Santa Monica. We found that it generally met the specifications

that we asked for. It seemed to be very insensitive to fiber alignment

and things of that nature and we found that it was really a reasonable

system to use.

Steve Levy (U. S. Army Electronics & Technology Laboratory).

My question to the first two authors regards both transmitter and

receiver power supply in terms of noise generated in the supplies by

the EMP environment. What affect did it have on the response of

their optical transmitters and receivers, and on their logic?

Lyons . The power supply ripple and noise was very well filtered

before it reached the transmitters or the receivers. In the ground

base system we had to take much more care to make it quiet. The

transmitters are the main problem in that the diode is biased on all

of the time so any change in current or voltage is going to affect

the system. I don't believe we saw any effect in regards to EMP or from induced noise.

Levy . I am concerned about the fact that both the stability of the

power supply as well as the effect of EMP on the power supply itself.

We found that, even though we tried to protect our power supplies in our transient voltage monitors, we had to go to battery operation.

That solved our particular problem. Also, TTL logic is very sensitive

to noise along the power supply line. Could you analyze the origin

of the noise in the receiver end that gave you the problem between

the oscilloscope and your optical receiver?

321
John Stewart (Harry Diamond Laboratory). Are you speaking of the

EMP signal Stan described? We are still working on that problem by

going through a process of trying to triple shield and build appropriate

shield boxes for the actual component. We are able to live with the

EMP induced signal. We are able to identify the operating parameters

from the signal that we are getting now.

Mario Pi Capua (Physics International). I should have asked this

question back at the office, Stan. But what was your bandwidth of

your system?

Lyons . We specified initially 200 megahertz bandwidth at the

3 dB points. It turned out that the best that Meret could give us was 10 kilohertz to 130 megahertz at the 10 dB point and a usable

signal although it was down about 1 5 dB at 200 megahertz. The system

dynamic range at 130 megahertz ws about 45 dB.

Bob Kolibas (Raytheon Company). A follow up on that same question.

What was the distance in the length of the cable and did that affect

your bandwidth capability?

Lyons. The length of the cable is 100 meters and the bandwidth

capability depends upon whether you use the step index fiber or the

graded index fiber. We used the system that Meret suggested which

was the step index fiber and it gave us the best overall bandwidth.

It turned out that it was flatter over the range than we wanted to

use. The graded index had a larger slope on the signal-to-noise

than the dynamic range curves.

Rich ard Malewski (IREQ). You have been measuring voltage and current

from the EMP simulator. But you probably also recorded the

electromagnetic fields at the ground level nearby the test object

with some sort of probe. What is the requirement on the bandwidth of

the probes and on your voltage and current monitors? Did you use the

same optical fiber system for signal transmission for the probes and

for the monitoring devices for voltage and current?

Stewart . We use a differential voltage probe or current probe,

dependi ng on the requirement. But we use a different fiber optics

data-link system. It's a 1 50-megahertz bandwidth system with remote

control attenuators which was developed by Harry Diamond Laboratory.

The fiber optics that are used to generate the signals to the operating

system of AESOP are a little different than the data acquisition

data-link systems.

Hebner. Before we get off on this, I want to get one further

clarification. To my earlier question you answered that you estimated

about a 5 percent linearity accuracy. I would like a little further

explanation of this. Some of the original voltage waveforms, that I saw very early in the first presentation, had oscillations on them

which seemed to come from at least two or three different sources.

322
I estimated the magnitude of these oscillations to be between

10 and 20 percent of the peak voltage. Later, I assumed that

those resulted from the leakage problems which you talked about

but now I realize that those may be real waveforms that result

from reflections in your system. Are you measuring those signals to

within 5 percent accuracy and are you working at trying to improve

the accuracy below 5 percent? If you are, I think that it is remarkable

Lyons . The high frequency ringing you saw on those waveforms was the

EMP leakage into the system. We ran the system at lower levels so that the EMP generated by the simulator was not as great an amplitude

and that is where we compared the down link transmitter waveforms

with the on-board diagnostic waveforms. We determined the 5 percent

linearity of the waveform by examining the frequency content of the

waveform that we were interested in. As we were saying, the noise is

still a problem and I don't think that there is anyway that we know

of right now to solve that problem completely.

Hebner . By linearity, do you mean the peak voltage as measured?

Would you define linearity?

Lyons . OK. I guess maybe I should say waveform fidelity instead.

If the pictures were normalized in amplitude and overlayed the

perturbations on the waveform were within 5 percent at all points.

Hebner . How did you measure the voltage on the high side to 5 percent?

You are comparing it with something else, and these voltage levels of

5 percent measurements are fairly impressive. I was just wondering

what device you are comparing your fiber optical system against.

Lyons . The standard was putting the scope in the screen box on board

the system and measuring that same waveform without sending it down

to the fiber optic system. Again, we were more interested in waveform

fidelity than the absolute voltage value.

Hebner . Thank you.

Boyer . Are there any further questions? I have one on the Los Alamos

paper. You mentioned early in the paper that you have some sort of

monitor to detect the faults in high voltage cables. I was wondering

if you would expand on that a little bit.

• Bill Scarlett (Los Alamos). What we have is a B loop in a high

voltage feedthrough insulator. We look for the direction of current

flowing in each of six cables feeding each anode. Since the cables

are shorted together at the anode, a fault in one of those cables

will lead to current flowing in the reverse direction. This leads to

an anomaly in the timing of the B signal for the failed cable.

Levy. I have a question on the Antares paper. You mentioned the

323
fact that there was a problem of lining up the transmitter fiber

optics and in the paper you also mentioned that you had a pigtail

lead on the optical transmitter. That means to me then that there

was a problem of splicing the two optical fibers, one coming out of

the pigtail to the one going into the line. If this is so, could you

please explain the problem in a little more detail because this is

certainly going to be a fundamental problem that everyone is going to

run into if we do go into optical transmission of data. If this is

not so, maybe you could explain where the source of error is coming from.

Scarl ett . The main source of error is the centering problem in the

fiber optic connectors. We measure and transmit a very large number

of signals as was indicated on the viewgraph. There are on the order

of 100 pulsed quantities and well over 100 timing channels. We use similar types of fibers and terminations in our control channels, so the total number of channels is large. Because of cost considerations,

we picked the plastic and connectors, which are one of the cheaper

standard connectors available. There are other connectors in which

the centering problem has been solved by fairly elaborate means. Unfortunately, these connectors are also fairly expensive. The

problem is there, but it can be solved. The main problem occurs with

two fibers of similar size.

324
DATA RECORDING TECHNIQUES FOR THE

SANDIA PARTICLE BEAM FUSION ACCELERATOR

William B. Bayer and Eugene L. Neau

Sandia National Laboratories

Albuquerque, New Mexico 87185

ABSTRACT

A data handling facility has been developed for the Sandia

Particle Beam Fusion Accelerator (PBFA-I). The facility records and

processes nanosecond diagnostic signals from both machine performance and experiment diagnostic monitors. The system consists of an

EMP-shielded cable plant, two minicomputers, coax switches, transient

waveform recorders, CAMAC pulse processing modules, and trigger/fiducial timing generators . All components are calibrated under either

manual or computer control. Waveforms are recorded and calibrated

by the system with an amplitude accuracy of + 1%. Signals frcm

different monitors can be aligned in time to within + .4 ns. The

facility is capable of supporting multiple simultaneous users. Applications software has been developed to support calibration,

data acquisition, and data reduction. The facility is operational

and is supporting two accelerators.

1 . Introduction

This paper describes an on-line data handling facility for

Sandia' s Particle Beam Fusion Accelerator, PBFA-I, and the upgrade

prototype machine SuperMite. These accelerators are used for research

on inertial confinement fusion (ICF) using particle beams. The main

main objectives in designing the data acquisition system were:

1. process both experiment and machine performance diagnostic

signals,

2. record high signal-to-noise ratio, wideband waveforms in a

severe EMP environment,

3. support multiple users recording and analyzing data

simultaneous ly, and

4. provide fast turnaround for experimental results.

325
Commercially available equipment is used wherever possible.

However, several special purpose devices were developed. This data

handling facility is a significant upgrade of an existing system that

supports other Sandia particle beam fusion research accelerators. [1]

2 . Accelerators

PBFA-I has 36 independent beam lines and is capable of delivering

30 terawatts and 1 megajoule in a 2 to 4 megavolt, 35 ns pulse into a

target chamber. [2] The accelerator can be configured to generate

either electrons or ions. The objective is to deliver sufficient

beam energy to demonstrate the scientific feasibility of ICF. The

accelerator upgrade, PBFA-I I, will provide 3 times as much power in

the 1985 time frame. The SuperMite accelerator is a 4 module prototype used to develop higher power components. Other small accelerarators are also planned for the facility.

3. Cable System

Signals to be recorded are fast transient pulses of 10-1000 ns duration and 1-5000 volt peak amplitude. These waveforms must be

recorded in the presence of EMP levels generated by di/dt and dv/dt

of 10-^ amps and volts per second. This EMP level requires that

signal cables be carefully shielded over the entire run. A picture

of the cable plant is shown in Figure 1.

Figure 1. Cutaway view of PBFA-I with fixed cable plant.

326
Jumper cables run from diagnostic monitors to feedthru ports near

the floor of the tank. These consist of RG-214 cable run inside

flexible solid convolute shield tubing. Special adapters were

designed to mate the shield conduit to type N and HN connectors in

an electrically and mechanically sound fashion. Ferrite sleeves are

inserted in the adapters to further reduce noise pickup. Figure 2

shows an assembled cable and the convoluted shield. The entire

I-beam support structure beneath the tank was made rf-tight. This

enclosure contains 800 RG-331 cables which run from feedthru ports

in the bottom of the accelerator to a shielded splice room. All

cables serving similar diagnostics were cut to equal lengths to

+ .5 ns. From this point signals can either be recorded and processed

locally or jumpered to the main screen room for processing. There

are 270 RG-331 cables run from the splice room to the screen room in

nine 6-inch diameter conduits. A tenth conduit is available for

further expansion. Unshielded RG-214 cables are used as jumpers in

both the splice and screen rooms.

Figure 2. Shielded jumper cable connector assembly and stripped back

convoluted shield.

4. Computer Hardware

Most data recording and processing electronics are housed in a high quality double-shielded rf screen room. A photograph is shown

in Figure 3. This room is 15 'W x 65 'L x 10' H with a raised floor.

327
Figure 3. Data acquisition facility main screen room hardware.

A block diagram of the waveform recording system is shown in

Figure 4. The main computer is a Hewlett-Packard HP-1000 system

with 256 Kb of 16 bit memory and a hardware floating point processor.

Mass storage consists of a 20 Mb moving head disc drive with a 10 Mb

removable cartridge and two 256 Kb floppy disc drives. A nine track,

45 ips, 800 bpi tape drive is also available. The system console is

an HP-2648 graphics terminal with dual cassette tape drives. Two

Tektronix 4014 Display Terminals with Enhanced Graphics Modules are

available for data acquisition and reduction. Each terminal has a 4631 hard copier with 4-channel input multiplexer. A Versatec 1200A

printer/plotter is also available. This device has a Versatec

5-channel input multiplexer to enable the plotter to make high

quality hard copies from the Tektronix terminals. The data acquisition HP-1000 is tied via an HP DS-1000 intercomputer link to a similar computer in the PBFA facility control room.

5. Waveform Digitizing Hardware

The facility has 44 Tektronix 7912AD Transient Digitizers with

7A16P programmable vertical amplifiers and 7B90P programmable time

bases. The vertical amplifier has a specified minimum bandwidth of

200 MHz. This is adequate for recording virtually all accelerator

and experiment diagnostic signals.

328
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Figure 4. Block diagram of waveform recording system.

The 7912AD's are interfaced to the HP-1000 computer via IEEE 488

General Purpose Instrumentation Buses (GPIB) . Each GPIB is specified

to handle 15 devices, including the controller. However, the bus

would intermittently fail whenever more than eight 7912AD's were

powered up on that bus. This problem was solved by installing

capacitors on the HP controller cards to slow down the handshaking

process. The fastest pulse widths were increased from 300 ns to

600 ns. This enabled bus signal levels to reach a full 2.5V.

Data signals from up to three different experiments are cabled

to the input ports of a computer controlled, four-position coax

rotary switch module. Computer generated calibration signals are

connected to the fourth port. The coax switches are Tektronix SM-2

matrices with twelve modules per unit. The SM-2's are controlled

by 32 bit digital output cards in a Tektronix 1340 Data Coupler.

The Data Coupler also houses the vertical and horizontal reference

signal generator cards used for autocalibration. The Data Coupler

is interfaced to the HP-1000 computer via an RS-232 link operating

at 1200 baud.

The output signal from the coax switch is fed into a programmable

attenuator. A fiducial timing marker pulse can then be added to the

329
signal via a resistive power adder. This configuration requires

an isolation amplifier in the fiducial signal line to prevent crosstalk among data signals through the fiducial fanout network. In

some cases the fiducial signal is routed to the 7912AD vertical

amplifier B channel, and timing information is only recorded during

the preshot baseline check.

The programmable attenuators are Allen Industries Model 50DA63.

Attenuations of 0-63 dB may be selected in 1 dB steps by relay

closures. They are controlled by standard relay driver cards in

Hewlett-Packard Multiprogrammer chassis. The Multiprogrammers

are interfaced to the computer over the GPIB. The attenuators can

reliably withstand 500 V, 100 ns pulses. Higher energy signals are

routed through special fixed attenuators in front of the coax switch.

These devices consist of 3 and 8 W low inductance resistors mounted

on 50 Q stripline boards in a minibox with type N connectors.

Precision timing is required to record and analyze the diagnostic

signals. All cables have been built to + .5 ns as measured by a cable length digitizer. Special trigger and fiducial marker pulse

generators were developed for this facility.

The trigger generator was built by Science Applications, Inc.

(SAI).[3] The unit is fabricated as 6 NIM-type modules. Analog,

digital, manual, and internal rep rate inputs are provided. Each

analog input has a slope/level discriminator. Minimum delay time

is 30 ns for both analog and digital inputs. Outputs are divided

into four groups of 16 each. Each group is separately delayable

from nominal 0-10 fis in 10 ns steps. Asynchronous clocks are used

to yield a shot-to-shot jitter of < 1.0 ns. All functions of the

trigger generator can be computer controlled via the GPIB.

Programmable functions include; input enable for any or all channels;

analog slope and level; digital slope; and output delay.

The fiducial generator was also developed by SAI.[4] This

device will trigger on a fixed minimum input pulse level of 2 volts.

The unit has ten outputs of 35 volts simultaneous to + .1 ns. Output pulse width is variable from 2-20 ns in 2 ns steps. The

width may be increased by increasing the length of the internal 10 Q

printed circuit charge line.

The data system has the requirement to record almost 400 diagnostic signals to fully monitor PBFA-I performance. This precludes

using 7912AD's. Sandia is funding development of a multichannel

waveform digitizer to record transient signals. The design goals

for this device are 40 channels sampled simultaneously at intervals

variable from 1-50 ns and a record length of 240 samples per channel.

Full scale amplitude sensitivity is fixed at 2 V. A + 1 V offset is

available to allow efficient recording of of unipolar or bipolar

signals

.

330
A block diagram of the multichannel recorder is shown in

Figure 5. Input voltage signal channels are converted to light by

fast LED ' s . All light outputs are routed by fiber optics to the

photocathode input faceplate of a streak camera tube. The electron

beams from the photocathode are accelerated, focused, and swept

across the phosphor screen output faceplate. Fiber optic faceplates

are used on both input and output to maximize light transfer. The

light from the phosphor is carried by a fiber optic reducing bundle

to a commercial TV format Charge Coupled Device (CCD) chip. The CCD

serves as an analog storage device. Data signals are clocked out

from the CCD, digitized, and stored in a digital solid state memory

at conventional rates. The multichannel recorder is interfaced to

the computer via the GPIB. The initial prototype is scheduled for

completion in 1981.

6 . Software

The HP-1000 computer runs under the RTE-IVA operating system

with the Multi-Terminal Monitor. The only modification required

on the system software was to enable the GPIB driver to schedule a user program upon receipt of a service request interrupt (SRQ)

Previously the driver could only be enabled to auto-poll predefined

devices. However, the autopolling would not process extended

talker and listener devices which require secondary addressing.

This feature is used in the 7912AD's to distinguish mainframe,

vertical amplifier, and time base commands.

Applications software for the PBFA data handling system is

functionally similar to that developed for the previous facility.

However, all programs have been rewritten in a structured, tabledriven format to maximize maintainability and expandability. The

programs were also designed to be as machine independent as possible.

Approximately 22,000 lines of Fortran code have been developed.

Major programs support the following functions:

1. Automatic digitizer set up and data collection.

2. Automatic digitizer calibration.

3. Interactive data reduction.

4. Shot archiving.

Fortran was used because it was the only supported high level

language available for the HP-1000 when the project was started. All

hardware configuration parameters are placed in tables. Input table

data is in ASCII format and may be modified by the source editor.

Special programs translate the source tables to mixed-type-variable

records and store them in random access files. Special subroutines

access these tables for applications programs. Thus hardware capabilities may be modified and expanded without rewriting software.

331
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The automatic data acquisition program (AQD) begins by reading

and processing a table of signal channel parameters for a given

shot. The program uses this and calibration tables to set up all

signal recording equipment to be used for the shot. It then digitizes

a baseline- and checks each digitizer for proper operation. AQD then

arms for the shot. Finally, after the shot, the program reads,

calibrates and stores each waveform on the disc. The calibration

process consists of:

1. Averaging to center of trace,

2. Multiplying in vertical amplifier gain, attenuations, and

gauge scale factors,

3. Correcting for time base sweep nonlinearity,

4. Compensating high frequency cable losses.

After AQD has stored all the waveforms, they are available for

user processing. An Interactive Data Reduction (IDR) program is

used extensively for manipulating and plotting signals. This program

provides the following capabilities:

1. Interactive or procedure file macro-type command execution.

2. Access data from multiple shots on either hard or floppy

discs.

3. Array arithmetic operations — add, multiply, divide,

square root, exponentiation, raise-to-power, minimum,

maximum, average, integration, differentiation.

4. Fourier transform, convolution, deconvolution, low pass

filter.

5. Single, multiple, overlayed, and windowed plots.

6. Cursor control and printout.

7. Control of all programmable hardware functions.

Figure 6 shows an example of a waveform processed using IDR

commands.

An automatic calibration program (ACL) is used to generate and

maintain 7912AD and programmable attenuator calibration data. This

program generates cal signals and switches them to the appropriate

digitizer. It digitizes the resulting waveforms and computes calibration factors. These data are stored on disc for use by programs

AQD and IDR in calibrating signal waveforms.

Accelerator shots are archived on floppy discs. This method is

superior to mag tape in that it provides rapid direct access to old

shots. The following information is stored for each shot:

Shot Set Up Header

Calibration Data

Baseline Levels

Calibrated and Uncalibrated Cata Records

Cable Compensation Functions

333
Figure 6. Data signals processed by program IDR. Vacuum interface

di/dt, integral with respect to time and magnitude of

Fourier transform of each in dB.
7. System Performance

The PBFA-I waveform recording system is a complex combination

of hardware and software components. Tests have been performed to

determine the overal amplitude and time accuracy of the system.

Calibration techniques for the various components that affect the

signal amplitude are described below.

The 7A16P vertical amplifiers are computer calibrated byinjecting a single known dc voltage into the input and measuring the

deflection. This process is averaged over four tries. Each of the

six sections in the programmable attenuators is also computer

calibrated by injecting known dc voltages into the unit and measuring

output with the associated transient recorder. Fixed attenuators

are manually calibrated using dc power supplies and voltmeters. All

dc equipment is corrected to + .1% using a calibrated Fluke 382A dc

voltage reference source.

Cable runs are calibrated by a hybrid computer/manual technique.

A fast rise step pulse is injected directly into a digitizer and

recorded. The same pulse is then injected into the cable run to be

calibrated and recorded by the transient recorder. A constrained

deconvolution process is used to compute a compensating impulse

response function for the cable run.

The automatic data acquisition program uses the calibration

factors described above to process each recorded waveform. The

overall accuracy of this system was checked as follows. A 150V,

50 ns rectangular pulse was recorded by a transient digitizer. This

signal was then manually calibrated using .deflection factors determined by the 382A reference source. The same pulse was injected

into the input of a cable run. It was recorded and processed by the

data acquisition program: A least squares comparison was made of

the two waveforms. The overall scale error was .4% and the standard

deviation normalized to the peak level was 2.8%. The undegraded

signal, recorded degraded signal, and calibrated signal are shown in

Figure 7. An overlay plot is shown in the fourth quadrant.

The 7B90P time bases are calibrated by the computer. Periodic

waveforms with known frequency are recorded by the 7 9 12AD' s. The

software locates zero crossings on the waveform and generates a

calibration curve of time vs. zero crossing addresses. The data

acquisition program uses this curve to process recorded signals

.

The curve is assumed to be nonlinear. The time at which each sample

occurs is computed by a double sliding parabolic interpolation. [5]

The signal data points are transformed into an array of data points

occurring at a fixed sample rate by linear interpolation.

Accurate alignment in time of signals recorded on different

digitizers is critical for analyzing accelerator and experiment

performance. Alignment information is recorded by placing a narrow

fiducial timing marker on each trace when a baseline is taken.

335
Fig. 7. System amplitude and cable compensation test waveforms:

(a) undegraded pulse, (b) recorded pulse, (c) calibrated

pulse, and (d) overlay of (a) and (c).

336
The data acquisition software locates the peak of the fiducial pulse.

Ideally the fiducial pulse would toe added onto the data signal as it

is recorded. However, the lack of a broadband isolating power adding

technique precludes this at this time. The only error in taking the

fidu location from the baseline trace is caused by jitter in the

time base which is specified at .1 ns for a .4 ns risetime trigger

pulse.

After each signal is time calibrated, the leading edge is

truncated at a time equal to the distance to the fidu peak plus

one-half of a division. This truncation aligns all signals in time.

Tests have been made to determine accuracy of the data recording

system in aligning signals. A pulse generator with ten 20 ns outputs

simultaneous to within .1 ns was used. The pulses were injected

into the input, of the cable run and recorded by the 7912AD ' s at

10 ns/div. The waveforms were processed by the data acquisition

software. This process was repeated five times to separate fixed

time delays from shot-to-shot jitters. Fixed time delays are

caused by unequal cable lengths.

Least squares comparison fits were made of different signals

over the same shot. This data was averaged over all shots to get

fixed time delays. The fixed time delays were removed and the data

was analyzed for different signals over the same shot and for the

same signal over different shots. The results are that the expected

alignment error for a given shot was .26 ns with a worst case spread

of .96 ns. The expected shot-to-shot alignment error was .33 ns with a worst case spread of 1.04 ns.

The most significant source of error is the software process of

locating the peak of the fiducial timing marker pulse. Digitization

noise results in an expected error of about +1.5 sample. At 10 ns div ( .2ns/sample) , this is + .3 ns.

8. Accelerator Performance Monitors

Many accelerator performance diagnostic signals may be adequately

monitored by a single parameter once the machine operation has been

characterized. These parameters include switch closure times, peak

amplitudes, and time integrals. A stand-alone accelerator Performance

Measuring and Evaluation System (PMES) is being developed to record

and process single point data. A typical signal channel is shown in

Figure 8.

Commercial CAMAC modules are used to record approximately 300

channels of signal parameters. Special NIM-type signal conditioning

and trigger modules have been developed to support this systen.

Inputs to CAMAC time digitizers are buffered by 16-channel avalanche

transistor discriminator modules. Inputs to CAMAC gated peak

detectors and integrators are processed by modules which can contain

337
any coribination of attenuators, power splitters, pulse inverters,

and integrators. Trigger fanout modules with four TTL input channels

and six low level NIM outputs per channel are used to start time

digitizers. Gate generator modules with one TTL input and four

low level NIM outputs are used to drive gated peak detectors and

integrators. Commercial NIM time delay generators are also used

in the system.

INTERMEDIATE STORE TRIG- V-DOT

,—, SIGNAL CONDITIONING

GATE GENERATOR

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HP- 1 000

COMPUTER

Figure 8. Data channel hardware for recording intermediate store

V-dot peak amplitude and gas switch firing time.

The PMES operates in a stand-alone mode with a Standard

Engineering MIK-ll/2 crate controller with dual double-density

floppy discs. The CAMAC system will be located in the cable splice

room below the accelerator tank. This location was chosen because

of the limited number of cables running to the main screen room. The MIK-11/2 is interfaced to the main facility CPU via an RS-232

fiber optic link.

Software for the PMES system is functionally similar to that

for the HP computer. Many routines were directly transportable.

Machine dependent subroutines were rewritten. The relatively limited

capabilities of the PMES computer forced some simplifications in

the software system design. Programs were written to support

automatic data acquisition, interactive hardware manipulation,

interactive calibration, and transfer of data to and from the HP

computer. Programs have been written for the HP computer to enable

interactive and automatic data transfers between the two systems.

PMES is currently monitoring the signals shown in Table I. The

Marx generator time digitizers are normally triggered by a signal

from the Marx pulser unit (MPU) trigger. A second trigger pulse is

created by adding signals from all the Marx voltage monitors. This

pulse will trigger the time digitizers if one of the Marx generators

prefires before the MPU's are triggered. The Marx that prefired can

338
be located by noting which of the Marx time digitizers did not receive

a stop pulse.

Table I. Accelerator Parameters Measured with PMES

Monitor Point Parameter Number channels

Marx pulser unit Start time 8

Marx current Start time 36

Marx current Peak amplitude 36

• Intermediate store V Gas switch trigger time 36

• Intermediate store V Peak amplitude 36

• Line 1 V Peak amplitude 36

9 . Conclusion

The data acquisition and handling system described above is

operational and supporting PBFA-I and SuperMite . The 7912AD waveform

recording hardware and software have been running since January 1980.

The PMES system is currently being brought on line as PBFA-I is

characterized. The same system will also support other small

accelerators which are planned for the facility.

10 . References

[1] W. B. Boyer, "Data Acquisition and Processing on Electron Beam

Fusion Accelerators", IEEE Trans. Nucl. Sci., NS-25 , No. 1,

501-505 (1978).

[2] G. Yonas, "Pulsed Power Particle Beam Fusion Research", IAEA

Technical Committee on Inertial Confinement Fusion, Osaka,

Japan, Nov. 1979.

[3] K. R. Sites, A Modular Trigger Generator Unit Operations Manual , Science Applications, Inc., Tech. Man. SAI-81-26-LV, Las Vegas,

1980.

[4] K. R. Sites, SG-111 Fiducial Marker Generator, Science

Applications, Inc., Tech. Man. SAI-80-23-LV, Las Vegas, 1979.

339
[5] M. E. Bauder, Operations and Maintenance Documentation for

Program PLBRD , Sandia TC-TM-70-114, 1970.

12. Sponsor Acknowledgments

This work was supported by the U. S. Department of Energy under

contract DE-AC04-76-DP00789

.

340
AN EVALUATION OF DIGITAL RECORDING EQUIPMENT AND

NUMERICAL CORRECTION TECHNIQUES IN IMPULSE MEASUREMENT

R. Malewski,

Hydro Quebec Research Institute,

VARENNES, Quebec, Canada, JOL 2PO

T.R. McComb and M.M.C. Collins,

National Research Council of Canada,

Division of Electrical Engineering,

OTTAWA, Ontario, Canada, K1A 0R8

The application of analog-to-digital converters to the measurement

of short-duration impulses is discussed. The desired accuracy of the

measurement is used to derive the performance requirements of the analogto-digital converter. The practical application of this analysis is

illustrated by a brief summary of an inter-laboratory comparison of two

recorders using both full and chopped lightning impulses with crest

values approximately 100 kV. One recorder was a micro-processor con- trolled digital waveform recorder mounted in a special shielding enclosure developed at the Hydro Quebec Research Institute (IREQ); the other

was a National Research Council of Canada (NRCC) oscilloscope whose

performance had previously been demonstrated in work at NRCC on high

voltage impulse measuring standards. The relative accuracy of the two

recorders was measured and their ability to resolve glitches on a smooth

impulse was investigated. In order to investigate the ability of the

IREQ digitizer to correct for errors caused by the response time, com- parative measurements were made using two very different impulse voltage dividers connected in parallel. The faster divider was connected

to the NRCC oscilloscope and the slower one to the IREQ digitizer. The

digital processing capability of the IREQ system was used to calculate

an approximation to the original impulse and this reconstructed impulse

was compared with the measurement on the faster system. In addition

Fourier Analysis was used to reconstruct the impulse and the two

methods of reconstruction were compared.

Key Words: impulse measurements; analog- to-digital converters;

numerical correction; error caused by response time.

1 . Introducti on

Over the past decade considerable work has been done on the adaptation of digital recorders to high voltage environments. A microprocessor controlled waveform recorder, suitable for use in high voltage laboratories, has been developed at IREQ [1]. While this instru341
ment has been carefully evaluated there are presently no standards

governing the use and performance of digital waveform recorders. As a contribution to the development of such a standard, it was decided to

make comparative measurements using the IREQ digital recorder and an impulse oscilloscope developed at NRCC [2]. This oscilloscope's performance had previously been demonstrated in work at NRCC on high voltage measurement standards [3,4]. The relative accuracy of the two sets

of recording equipment was measured for full and chopped lightning

impulses

.

Since digital recorders operate by sampling data rather than by

continuously recording, doubt has been expressed to to whether fast

glitches (such as might be caused by a partial breakdown in a transformer under test) will be recorded. The relative ability of the two

sets of recording equipment to resolve glitches on a smooth impulse was investigated using the partial breakdown of an insulator string.

A major advantage of digital recorders is that the record of the

impulse is available as a digital file and can be easily processed by

computer. It is theoretically possible to reconstruct the input to an impulse measuring system provided that both the output and the step re- sponse of the system are known. In practice some information is destroyed and for a sufficiently poor impulse measuring system reconstruction

is impossible. However, it is possible that there may be impulse measuring systems which are not sufficiently accurate for a particular type

of impulse but whose responses could be corrected by the application of

Duhamel's Theorem. Work on evaluating this possibility has been started.

2. The Application of Digital Recorders to Impulse Measurements

The use of analog-to-digital converters (ADC) in impulse measurements introduces errors because the signal is sampled at discrete points

and these errors can be compounded by the limited resolution of the con- verter. Quantization errors, aperture time and aliasing contribute to

the overall error.

Quantization errors of ± \ least significant bit are caused by

approximating the analog signal by a series of discrete values each of

which remains constant for at least one sampling period. This quantization error introduces high frequency noise as the ADC shifts levels

abruptly. Care should be taken to ensure that this noise is not accidentally increased during numerical processing of the recorded digital

signal

.

The time during which the signal is acquired (aperture time) should

be short compared with the sampling period since practical ADC cannot

sample sufficiently quickly to make use of their full resolution when

sampling fast signals.

If the frequency spectrum of the input impulse contains components

of significant amplitude at frequencies close to half the sampling rate

342
then aliasing may occur. The frequency spectrum of the input pulse,

the sampling rate and the accuracy of the ADC determine whether the

error caused by aliasing is significant.

When the rate of change of the input signal is sufficiently low

then the signal level can lie half-way between two adjacent digital

levels for several or many sample intervals. When this occurs the ADC

may change from one level to the next and back again arbitrarily gener- ating an artificial glitch. An example of this effect is shown in

Figure 1. Such glitches can be removed by software filtering but this

reduces the response of the system to real glitches [1].

FIGURE I. FALSE TRANSITION ERRORS ( f ) When the rate of change of the input signal is sufficiently high

most ADC will be subject to skip code errors [5,6] since the least significant bit (LSB) is unable to change quickly enough. For example when

an 8 bit ADC is recording a 10 MHz sine wave, whose amplitude is half the

full scale deflection, the LSB has to change at approximately 1 GHz.

There is presently no generally accepted method of specifying the

high frequency performance of ADC. One method in use considers the

response of the ADC in terms of the sampling error involved in measuring

the rate of change of a high frequency sine wave at its zero crossing.

This method emphasises the importance of the aperture uncertainty. A more recently developed method is to compare the measured rms error of

the digitized high frequency sine wave with the calculated rms noise

for an ideal ADC. While these methods are useful for the comparison of

various ADC they are not directly applicable to impulse measurements.

Elementary considerations require that if the crest value of an impulse is to be measured to within x% then at least one sample point

must fall within the time interval for which the impulse is expected to

343
be within x% of its crest value. Similarly for a 30 to 90% risetime

(or 10 to 90%) to be measured accurately sample points should occur sufficiently close to times of occurrence of these values for interpolation errors to be negligible. The application of these requirements

to the measurement of a standard lightning impulse (front time 1.2 ys,

time to half value 50 ys) is illustrated in Table 1 in terms of the

sampling rate needed to fully utilize the resolution of the converter

for measurement of crest voltages and the rate needed to measure the

front time (30-90% amplitude) within 10%. When there are oscillations on the impulse a much higher number of samples will be needed to define the

oscillations. When the shape of the pulse is known or when numerical

correction is used the required number of sample points may be smaller.

TABLE 1 REQUIRED SAMPLING RATES

No. of

Bits

8

10

12

Resol ution

of Amplitude

Measurement

% Full Scale

0.4

0.1

0.025

Required Sample

Rate For Crest

Measurements

(MHz)

34

7 20

Required Sample

Rate For Front

Time Measurements

(MHz)

14

14

14

3. Inter-Laboratory Comparison of Recorders

3.1 Equipment

The I REQ recorder [1] consisted of two input channels, a fast ADC

and memory, and a graphic computer terminal, all housed in an electromagnetic shield and supplied with AC power through RF filters and an isolating transformer. Each input channel consisted of a 1 Ml] input

impedance attenuator followed by a fast buffer circuit to match the

attenuator output to the 50 £2 input impedance of the ADC. The ADC had

a resolution of 8 bits, a full scale deflection (fsd) of 10 V on its

range and a maximum sampling frequency of 10 ns per

channel input at the maximum sampling frequency was used

work and each waveform was initially stored in the 2

These records were then copied to the intelligent

they were scaled and displayed with appropriate parameters

least sensitive

sample. Single

throughout this

kilobyte memory,

terminal where

The amplitude resolution of the instrument was 1 in 256 or approximately

0.5% fsd which resulted in a maximum possible reading error in amplitude measurements of 0.8% fsd. The corresponding figures for the time

measurements were 1 in 2048 or 0.05% and 0.1%.

The NRCC oscilloscope was a Southern Instrument transient recorder

circa 1960 which had been extensively modified at NRCC to improve its

shielding and linearity [7]. The bandwidth was DC to 300 MHz (-3 dB)

and the fsd was 1000 V for signals input directly to the deflection

plates. Each impulse was photographed and the record was read on an

344
optical comparator with a resolution of 2.54 um (0.0001 inches) corres- ponding to a resolution of at least 0.2% for the deflections recorded

in this work. The relative accuracy of this type of oscilloscope has

been measured by comparing records from two oscilloscopes connected in

parallel to one divider: the difference in readings of crest voltage

was 0.44 ± 0.5% (with a 95% confidence interval [8]).

Full and chopped lightning impulses were generated using the

circuit shown in Figure 2. The three dividers used in this work were

screened resistor dividers which were rated for full lightning impulses

with crest values of 900 kV, 350 kV and 100 kV; details of the dividers

used are listed in Table 2. Experimental response times of the impulse

measuring systems were determined using a low voltage dc power supply

and a mercury-wetted relay [3,4]; this relay was approximately the same height as the chopping gap and so no additional vertical lead was required. An example of an experimental unit step response for a three

component system is shown in Figure 3. The response times of the three

component systems were measured using resistors from 100 Q to 900 ft and

the response times of two component systems were determined by extrapolating the zero resistance. (Table 3).

i so a Wv

2.5 m STAGES

SI X RESISTOR

DIVIDER

OF MARX

GENERATO R

FIG. TWO — CIRCUI T FOR LIGHTNING I M P U L S E S. D E T A I L S OF

THE IMPULSE MEASURING SYSTEMS AND CHOPPING

GAPS ARE GIVEN IN THE TEXT.

TABLE 2 PHYSICAL PROPERTIES OF THE RESISTOR DIVIDERS USED

Rated Withstand

Voltage (kV)*

Insulation Resi stance

(kfi)

Height

(m)

900

350

ai r

ai r compressed

nitrogen

30

10

2.7

1.0

100 10 0.13

* for full lightning impulses

345
.6 -

.2 -

".2 i i i i i i i i i 0 10 20 30 40 50

t(ns)

FIG. THREE — UNIT STEP RESPONSE OF

THE PRESSURIZED DIVIDER INATHREE

COMPONENT SYSTEM ; DAMPING RESISTOR

434 OHM, RESPONSE TIME 2.2 NS

TABLE 3 SYSTEM RESPONSE TIMES

Di vi der Dampi ng Response Time Calculated Response Times With

Res i s tor (T with 500 NRCC Oscilloscope IREQ Digitizer

MHz Digitizer)

(ohm) (ns) (ns) (ns)

900 kV 877 105* 105 112

900 kV 0 55 55 68

350 kV 434 13* 13 42

350 kV 0 2.0 2.2 40

100 kV 434 2.2* 2.4 40

100 kV 0 -4.1* 40

* Measured Val ues

3.2 Relative Accuracy of Recorders

The two recorders were connected in parallel so that impulses could

be applied to both recorders simultaneously. The NRCC oscilloscope was connected through 30 m of RG17 coaxial cable to the base of the high

voltage arm of a resistor divider. The low voltage arm was mounted at

the NRCC oscilloscope. The IREQ digitizer, whose input impedance was

1 Mfi in parallel with approximately 50 pF, was connected to the input of

the NRCC oscilloscope through 2 m of RG8 cable. Full and chopped

lightning impulses were applied to the impulse measuring system.

The results of the comparative measurements are given in Table 4

for full lightning impulses and in Table 5 for chopped lightning impulses.

346
TABLE 4 MEASURED VALUES OF THE CREST VOLTAGES OF

FULL LIGHTNING IMPULSES

(Same Input to Both Recorders)

LJ f\ /» /~\ \,rt /-i Kecora 1NKLL Va 1 Ue 1 KHj Value A*

Number (kVj [KM) 01

To

5 50.6 50.9 +0.6

5A 50.8 50.9 +0.2

8 50.8 51.5 +1 .4

9 65.4 66.0 +0.9

10 66.6 66.0 -0.9

11 66.7 66.6 -0.2

403 101.9 102.1 +0.2

557 100.4 102.1 +1 .7

Average A 0.5

Standard Deviation 1.0

Maximum Possible Reading Error 0.6

IREQ Value - NRCC Value

^ NRCC TaTue '

TABLE 5 MEASURED- VALUES OF THE CREST VOLTAGE OF

CHOPPED LIGHTNING IMPULSES

(Same Input to Both Recorders)

Record NRCC Value

Number (kV)

22 49.4

23 48.7

24 48.3

25 49.2

64 88.0

400 94.0

402 97.5

501 48.6

503 48.9

IREQ Value - NRCC Value

NRCC Value

IREQ Value A*

(kV) %

50.6 +2.4

49.0 +0.6

47.1 -2.5

47.7 -3.0

86 . 0 -2.3

95.8 +1.9

97.9 +0.4

50.0 +2.8

50.0 +2.2

Average A 0.3

Standard Deviation 2.5

Maximum Possible Reading Error 1.6

Illustrative impulses are shown in Figures 4 and 5; on the IREQ re- cords the first 2 ys and the first 20 ys were plotted. The measurements of crest voltage of full impulses were within - 0.9, +1.7% and the

average difference was 0.5 ± 0.8% (95% confidence interval). These

results indicate that there was no systematic difference in the results

347
(a ) NRC OSCI LL06RAM

FIG. 4 — FULL LIGHTNING IMPULSE

t ( /i s )

( b ) I REQ RECORD

a) NRC OSCILLOGRAM

Fl 6. 5

t(/n)

(b) I REQ RECORD

CHOPPED LIGHTNING IMPULSE

from the two recorders, and that the differences in individual readings were within the maximum possible reading error. The measurements of

crest voltage of chopped impulses were within ± 3% and the average

difference was 0.3 ± 1.9% (95% confidence interval).

3.3 Glitch Tests

In order to evaluate the response of the IREQ digital recorder to a fast glitch, the partial breakdown of a transformer under test was simulated by using an insulator string, with one of the insulators paralleled

by a small spark gap, as a test object. When a full lightning impulse

of suitable amplitude is applied to this test object the spark gap breaks

down shorting one insulator and generating a transient. A string of six

insulators was used in this test and the spark gap was mounted on the

bottom one. The glitches generated in this test were approximately 3%

of the crest voltage. In all cases where a glitch was detected on the

oscilloscope a similar glitch was observed on the IREQ recorder. An

illustrative response is shown in Figure 6.

348
m

CTT1

(a) NRC OSCILLOGRAM

FIG. 6

t ( /A8)

(b) IREQ RECORD

GLITCH ON FULL LIGHTNING IMPULSE

4. Numerical Correction Techniques

4.1 Reconstruction of Impulses

Using Duhamel's Theorem (convolution theorem) the response, r(t),

of an impulse measuring system to an input, f(t), can be written in

terms of the step response of the system, g(t),:

r(t) = s\ f(t) ^ {g(t-x)} dx (1)

If r(t) and g(t) were known analytic functions then this integral equation could be solved to find f(t); unfortunately r(t) and g(t) are measured with only limited accuracy (± 0.5% of crest value in this work).

This inherent inaccuracy in any measuring system leads to a loss of information in the measuring system [9]. Whether this loss is significant

depends on the form of f(t) and g(t). One of the aims of high voltage

testing standards [3,4] is to set limits on the unit step response of

an impulse measuring system so that r(t) will be approximately equal to

f(t) (within the accuracy prescribed by the standard); the limits on g(t) depend on the form of f(t) and are prescribed for full and chopped

lightning impulses. The purpose of this work was to discover if an impulse measuring system which did not meet the requirements of the

standards could be used with numerical correction of the measured output.

Two correction techniques were used. One of these was the direct

numerical solution of the integral equation given in eq. (1) and the

other was the use of the Fast Fourier Transform (FFT) to transform eq.

(1) to the frequency domain.

4.2 Validation of Numerical Correction Techniques

In order to investigate the accuracy of numerical corrections for errors caused by response time, comparative measurements were made using

two very different impulse measuring systems connected in parallel. The

faster system (using the 100 kV divider) was connected to the NRCC

oscilloscope and the slower system (using the 900 kV divider) was con349
nected to the IREQ digitizer; the planes of the two impulse measuring

systems were at right angles to minimize coupling. Simultaneous measurements were made of impulses chopped on the front and numerical corrections were made to the impulse measured on the slower system. The re- constructed impulse was compared with that measured on the faster system.

4.3 Numerical Solution of the Integral Equation

Recently it has been shown that eq. (1) can be solved by expansion

into a Taylor series and that f(t) can be expressed as an infinite sum of the derivatives of r(t) [1Q] . When expressed in terms of finite

differences and limited to 2m + 1 terms this sum can be written:

x

n

= Wn-m + k nyn

+ Wn+m

(2)

In practice it is found that (2m + 1) can be quite small (3 to 7) [6,11].

The coefficients, , are found from the normalised step response and

used to calculate f(t) from r(t). It has been found that the convergence

of the method can be improved by using a software filter to remove high

frequency noise from r(t).

4.4 Numerical Solution Using the Fast Fourier Transform

The step response of the slow system (900 kV divider) was recorded

using a Tektronix WP 2200 system and the impulse response was derived by

three point differentiation using Tektronix SPS BASIC Signal Processing

Package. The program DCONVL from the Tektronix Software library was used to generate the input waveform by transforming the impulse measured

on the slow system and the impulse response into the frequency domain

and taking the quotient. This quotient is then transformed back to the

time domain to give the reconstructed impulse. The division process will

emphasize the effect of high frequency noise and may result in an un- realistic estimate of the input pulse. To reduce the effects of high

frequency noise on the reconstruction process a cosine taper is applied

to both the measured pulse and the impulse response. This taper can be

applied to the edges of the waveforms to smooth 10 to 50% of each edge.

In practice the minimum taper which gives convergence is used.

4.5 Estimations of Bounds on the Correction Term

Two simple methods can be used to give approximate correction terms.

Since the chopped impulses used had a lower rate of rise immediately

before the chop than at the mid-point (i.e. they were concave downwards)

aproximating the chopped impulse to a linearly rising pulse with a risetime equal to the virtual risetime [3,4] of the impulse gives an upper

bound to the necessary correction. If the convolution of the measured

impulse and the impulse response is taken the size of the correction

needed for a second pass through the measuring system can be calculated

and this operation is easily performed since it reduces the effects of

high frequency noise. This correction should always be less than or

350
equal to the correction needed for the effects of the first pass through

the system.

In practice the uncertainty of the measurement will appear in the

calculated corrections. The quoted corrections, which are based on the

measured values, are subject to ± 2.5% error from this source in addition

to any errors generated in the calculation.

4.6 Results of Numerical Corrections

It has been found that both methods of numerical correction over- estimate the correction needed. An illustrative result is given below:

Crest voltage measured on fast system 156 kV

Crest voltage measured on slow system 147 kV

Standard deviation of crest voltages 2.5 %

Measurement error caused by response time 6 %

Correction calculated from integral equation 13 %

Correction calculated from FFT 11 %

Estimated upper bound 13 %

Estimated lower bound 7 %

The impulse measured on the fast system is shown in Figure 7.

170 kv

ON THE FAST SYSTEM

351
The estimated upper bound for the correction needed to measure this

impulse is 0.2%. The measured and reconstructed impulses from the IREQ

system are given in Figure 8. In the FFT method, the reconstructed

impulse was convolved with the impulse response of the system and the

crest value of this calculated output was divided by the crest value of

the measured output to give the normalization factor. The measured output, reconstructed impulse and the calculated output (displaced in time) are shown in Figure 9.

Neither of the calculated input functions faithfully reproduce the

measured input. In particular the FFT method shows oscillations not

present on the input pulse. While these oscillations can be reduced by

increasing the filter weight or the % of taper applied no a priori

criterion has been developed for this. The criterion used in this work

was that the calculated output and the measured output pulses should

agree.

The measured error caused by response time is equal to the estimated

lower bound within the limits of the experimental error (± 2.5%). Since

both calculated corrections are close to the calculated upper bound it

is evident further improvements are needed to the correction techniques.

352
I 2

TIME

FIGURE 9. CALCULATED IMPULSES USING FFT

5. Conclusions and Future Work

Comparative measurements have shown that the IREQ digitizer was sufficiently accurate to meet the requirements of IEC Standard 60 - High

Voltage Test Techniques for the measurement of the crest voltage of full

lightning impulses and of the voltage at chopping of chopped lightning

impul ses

.

Measurements of glitches have shown that the IREQ digitizer's

sampling resolution was sufficient to resolve glitches caused by the

partial breakdown of insulation; an amplitude resolution of more than

8 bits would allow smaller glitches to be measured.

Initial measurements to evaluate techniques of numerical correction

of measured impulses have been made. These indicate that the techniques

used for correction overestimate the correction needed. It is planned

to continue this investigation and to develop both valid numerical

correction techniques and criteria to determine when numerical corrections can be applied.

353
6. References

[I] R. Malewski and A. Dechamplain, 'Digital Impulse Recorder for HighVoltage Laboratories'. IEEE Transactions on Power Apparatus and

Systems, Vol. PAS-99, pp. 636-649, March/April 1980.

[2] R. Malewski, T.R. McComb, and M.M.C. Collins, 'An Inter-Laboratory

Comparison of Digital and Analog Recording Equipment Used in Impulse

Measuring Systems'. Proceedings ofthe Canadian Communications and

Power Conference, Montreal, Quebec. October 1980, pp. 60-63.

[3] International Electrotechni cal Commission - Standard 60 - HighVoltage Test Techniques.

[4] IEEE Standard 4 - 1978 - IEEE Standard Techniques for High-Voltage

Testing.

[5] W.S. Zaengl , Discussion on Reference 1.

[6] R. Malewski, A. Dechamplain, Closure to Reference 1.

[7] F.C. Creed, 'An Improved Cathode-Ray Oscillograph for Impulse

Recording'. Bulletin of the Radio and Electrical Engineering

Division, National Research Council of Canada, Vol. 11, pp. 7-11,

Oct/Dec. 1961.

[8] International Organization for Standardization 2602-1973 'Statistical

Interpretation of Test Results - Estimation of the Mean-Confidence

Interval '

.

[9] International Research Group Renardieres on Impulse Measuring

Systems, 'Facing UHV Measuring Problems'. Electra No. 35, pp.

157-254.

[10] S.E. Kiersztyn, 'Numerical Correction of H.V. Impulse Deformed by

the Measuring System' . IEEE Transactions on Power Apparatus and

Systems, Vol. PAS-99, No. 5, Sept/Oct. 1980, pp. 1984-1995.

[II] R. Malewski, B. Langlois, Y. Gervais, 'Correction Des Erreurs de

Mesure des Chocs Rapides a H.T. par Traitment Numerique'.

Proceeding of the Canadian Communications and Power Conference,

Montreal, Quebec, October 1980, pp. 150-153.

354
ANALYSIS OF A POWER SYSTEM TRANSIENT RECORDING LABORATORY

Randall E. Poliner

Terry J. Reed

Westinghouse Electric Corporation

Advanced Systems Technology

Pittsburgh, PA 15235

Under a project sponsored by the Electric Power Research Institute,

two automated Transient Recording Laboratories have been developed to

measure and record naturally occurring electrical transient phenomena

on transmission lines. The laboratories are now installed and fully

operational on the Florida Power and Light transmission network at

500 kV and 138 kV substations. This paper describes noise suppression

techniques and methods utilized to insure data integrity in a high

voltage substation environment.

1. Introduction

The growing concern on the part of equipment manufacturers and

utilities as to the nature of transient surges on transmission systems

has necessitated the design and development of laboratories capable of

recording naturally occurring phenomena under actual operating

conditions. [1 ] To this point, most of the existing knowledge concerning

switching surges or fault conditions has been derived from staged tests or analytical studies. To supplement staged test data, the Electric

Power Research Institute contracted to produce an instrumentation system

to measure and record anomalous power system events with laboratory

accuracy and integrity.

2. Measurement Transducers

In order to obtain a complete description of each anomalous

occurrence relative to the instrumented transmission line, measurements

are made of five parameters on each of the three phases (Figure 1)[3].

These include

1) transmission line voltage

2) transmission line voltage as seen by the relay

3) transmission line current

4) transmission line current as seen by the relay

5) substation bus voltage

355
The transducers to measure these quantities are up to 300 meters

removed from the laboratory itself (Figure 2).

2.1 Transmission Line and Substation Bus Voltage Transducers

The line and bus voltage transducers are high accuracy, high bandwidth coupling capacitors. These CCPD's were modified to eliminate

the carrier frequency components and transformers, and install a 400

volt tap and capacitive divider network in order to condition the

voltage level for input to solid state electronics. The CCPD transducers have been shown in a high voltage laboratory environment to

have a bandwidth greater than 1.3 MHz while maintaining an accuracy of

better than 1%[3].

2.2 Relay Voltage Transducers

The voltage seen by the relay is monitored in the relay circuit at

the control house relay panel. Thus, prior to the laboratory sensor,

the signal has passed through the utility relay CCPD and been transmitted to the control house at baseband over nine conductor, Number 10

AWG control cable. At the 138 kV site, the control cable is unshielded,

229 meters in length, and the largest portion of the run is above

ground in a cable tray. At the 500 kV site, this control cable is

shielded, 305 meters in length, and run in a cable trough under the

substation ground grid.

2.3 Relay Current Transducers

As with the relay voltage monitoring system, the line current as seen by the relays is monitored in the relay circuit at the control

house relay panel. The utility's current transformers are multiratio

bushing type with ANSI C-800 ratings. At both sites, the line current

is composed of the sum of currents flowing through two circuit breakers,

each with an associated current transformer. The current flowing in

the transmission line is then the output of the two current transformers

on either side of the line termination connected in a differential mode.

The laboratory transducer located in the control house relay

panels monitors the utility current transformer secondary current. The sensor itself is a coaxial current shunt of approximately 0.01 ohms.

The shunt has been tested to assure a flat frequency response to

170 kHz.

2.4 Primary Line Current Transducers

The line current transducer is a free standing device developed

specifically for the program. The unique current transformer which

serves as the transducer features a special core design, winding

technique, and burden selection. The result is a transformer which

exhibits a flat frequency response (+ 0.3 dB) for 1 Hz to 100 kHz.

Significant saturation effects for current levels below 169 kA were also

356
FIGURE 1 - Transient Monitoring FIGURE 2 - Diagram of MeasureLaboratory installed at Florida ment Transducer Locations on Power and Light's 138 kV Lauderdale each of the Three Phases

Plant Substation

FIGURE 3 - Line Current Transducer

housing at line potential containing

current transformer, signal

conditioning, and signal transmission

electroni cs

FIGURE 4 - CCPD base housing

containing capacitor network,

signal conditioning, signal

transmission electronics, and

power supply

357
eliminated except for cases of successive, high magnitude fault currents

or transformer inrush currents.

The output signal of the current transformer is cabled to the

conditioning and signal transmission electronics, physically adjacent

to the transformer. The transformer, signal conditioning, and data

transmission electronics all float at line potential within a shielded

enclosure (Figure 3). The data transmission electronics are powered

by means of ultrasonic rod tapped at each end with piezo-electric

transducers and driven by a power supply providing a square wave voltage at approximately 20 kHz. In this way, the current monitoring

system is not dependent on, or affected by, the measured signal; thus,

the system is available to measure energization and de-energization

transients. The transformer and associated electronics are supported

by a nitrogen filled porcelain column which also contains the ultrasonic

rod and a fiber optic cable for data transmission.

3. Signal Transmission

The outputs of all primary transducers are conditioned to levels

compatible with solid state electronics, frequency modulated, and

converted to light signals for transmission over single glass fiber

optical cables to the recording laboratories. At the laboratories,

the light signals are reconverted to electrical signals, demodulated,

and cabled to the digital transient recorder (Figure 4).

3.1 Modulation and Demodulation

After conditioning, all signals are frequency modulated prior to

conversion to a light signal for transmission. The modulation and

demodulation are accomplished via state-of-the-art techniques in order

to assure data integrity, low losses, and high inherent signal to

noise ratio. Both circuits have been minimized to reduce crosssectional area and minimize lead lengths. Electromagnetic and radio

interference are further reduced through appropriate shielding

enclosures, conductive gasketing, and uniform copper ground mat.

3.2 Electrical/Light Conversion

Conversion of the frequency modulated electrical signal to an infrared light signal is accomplished by a light emitting diode, LED.

The LED operates with maximum spectral output at a wavelength of 825

nanometers with full radiated power of nominally 750 microwatts per

square centimeter at 150 milliamps. Coupling efficiency into the fiber

is approximately 50% due to the connector, numerical aperature of the

fiber, and uniformity of the radiant emittence as the distance from the

optical axis increases. Conversion from the light signal back to the

frequency modulated electrical signal is accomplished by a silicon

PIN photodiode. The photodiode features a typical dark current of 1 nanoamp and rise time of 1 nanosecond. The entire photodiode module

is matched to the transmitting LED and features a feedback network to

358
compensate for LED power drift, reduce noise susceptabi 1 i ty , and enhance

data integrity. Both the LED module and PIN photodiode modules have

been miniturized into 1" x 1/2" x 1/2" hermetically sealed cases. The

result is minimized Radio Interference and Electromagnetic Interference.

To eliminate capacitive loading and noise pickup, each module is

physically enclosed in the same case with the modulation or demodulation

circuitry, as appropriate.

3.3 Optical Fiber

The optical fiber used for data transmission is used for a variety

of purposes. In the case of the primary line current transducer, the

non-conducting fiber is used to carry the transmitted data signal from

the electronics at line potential to the laboratory at ground potential.

The optical fiber also provides for isolation from the various ground

potential shifts between transducers and the laboratories that occur

in the substation under transient conditions. The optical signals

are also immune to the intense electromagnetic noise found within a utility substation on a continuous and transient basis.

Two different types of optical fiber construction were used in

installing the laboratory. The first, ten mil fiber with hollow tube

construction, was found to be unsuitable for the utility environment

where great distances between equipment required enormous tensile

strength for pulling purposes. Another fiber of extruded construction

was found to have a higher actual tensile strength when terminated.

The fiber used is a single fiber, 8 mil, step index with attenuation

of 5 dB per kilometer at 900 nanometers. Terminations and splices

are accomplished through the use of conical, four rod connectors in

order to achieve high coupling ratios and low attenuation at splice

points. This splice and termination technique was also found to be

most suitable to field installation and repair.

4. Direct Noise Suppression Techniques

The utility substation environment is an extremely hostile EMI

and RI environment, particularly during transient conditions when

ground potential shifts between transducers and recording apparatus may

easily reach five kilovol ts. [2] Induced lightning and other impulse

type surges affect not only the ground plane, but also have a real

and measurable affect on the power system. Isolation, particularly

of low level electronic signals, can become an extremely complex

process. Isolation for the Transient Monitoring Laboratories is

primarily directed into five areas. These are the power supplies,

data transmission channel, grounding techniques, shielding techniques,

and laboratory enclosure. The data transmission channel and rationale

behind use of the optical fiber was discussed above.

359
4.1 Power Supplies

All power for the laboratory and associated electronics is derived

from a battery rack located in the respective site's control house and

a 60 Hz, 120 VAC inverter located in the laboratory. All power is

dispersed from the laboratory to the transducer electronics via

shielded cable. Power to each transducer is isolated through instrument

quality isolation transformers and dedicated DC power supplies. Power

to recording and other laboratory apparatus is also fed through

isolation transformers to reduce power supply induced noise from any one component to the other subsystems or from outside the laboratory

system. Power to the line current transducer electronics is, in the

final analysis, supplied through the use of an ultrasonic rod as discussed in Section 2.4.

4.2 Grounding Techniques

It was verified through the course of the project that the

inductance of ground leads could lead to high potential shifts during

transient conditions which would, in turn, severely affect the noise

performance of the system. For this reason, ground leads were kept as short as possible and as continuous as possible. Heavier apparatus,

including cabinetry, has been grounded through 20 mil by two inch

solid copper ribbon at several points. Circuit card ground planes

are accomplished by copper sheets where appropriate. Each enclosure

is also grounded with copper ribbon to bleed off extraneous radio

frequency signals. The improvement in observable noise due to the

strap grounding system was between 6 and 9 dB depending on the location

and frequency of interest. The improvement in signal-to-noise ratio

among the data channels was between 2 and 4 dB of coupled noise.

4.3 Shielding Techniques

Basically, each circuit has been shielded from the outside environment as well as each other. Enclosures designed to eliminate radio

interference are used to house each critical circuit. All enclosure

access covers are sealed, as required, with conductive gaskets. The

largest gain observed due to the shielding employed was seen in

reduced data channel cross-talk, especially at the demodulation site

where several data channels are adjacent to each other.

The laboratory itself is designed as an EMI free enclosure,

featuring double walls and copper gasketing around the door. The total

shielding system is capable of reducing the noise floor to -85 dB so that interference is insignificant compared to the specified 72 dB

signal-to-noise ratio.

5. Data Transmission Performance

An exhaustive series of tests were run with known inputs and known

power levels of white noise to precisely determine the data integrity

360
FIGURE 5 - Spectrum Analysis of

Transmitted Baseband Signal; Full

Scale Input with White Noise at

-50 dB

REF 15-0 dBn OFFSET 11 195.5 Hz

10 dB/DIV

1

1

\

RANGE 15.0 dB* -43.6 dB

START 40.0 Hz STOP 100 000.0 Hz RBH 1 KHz VBH 10 Hz ST 31-4 SEC

FIGURE 6 - Spectrum Analysis of

Received Baseband Signal

Using Input of Figure 5

361
FIGURE 7 - Spectrum Analysis of

transmitted baseband signal under

operating conditions

REF .0 dBv 19. 76 ?04.2 Hz 10 dB/OIV RANGE 2.0.|ifT- -71.7 dSv

CENTER 50 010. 0 Hz SPAN 99 980-0 Hz m 100 Hz VBH 300 Hz ST 20-0 SEC

FIGURE 8 - Spectrum Analysis of

received baseband signal under

operating conditions

362
loss in transmission. An example is shown in Figures 5 and 6 for a sample of tests run on a relay voltage channel. The results show approximately a 7 dB loss in the signal-to-noise ratio due to transmission

which is independent of the noise floor in the baseband.

Figures 7 and 8 demonstrate the actual performance of the

laboratory communication system under operating conditions. Figure 7 depicts the signal-to-noise ratio and dynamic range as observed at the

input to the FM transmitter, just after the transducer. Figure 8 shows

the demodulated analog signal in the laboratory. As can be seen, the

7 dB transmission loss observed above has been verified under

operational conditions.

6. Conclusions

The resulting field laboratory system is fully capable of monitoring and recording power system transient phenomena with unparalleled

data integrity compared to more conventional utility field testing

techniques [1,2]. The use of fiber optic based data transmission,

highly improved primary transducers, and elaborate grounding and

shielding techniques has yielded a measurement system with a dynamic

range greater than 66 dB and data of greater than 72 dB signal-to-noise

ratio. The importance of this high bandwidth data to the utility

industry becomes readily apparent in developing concepts and designs

for:

a) Compact lines, where phase-to-phase switching surges

are critical

.

b) New, nonlinear arresters, where the frequency and magnitude

of overvoltages become important.

c) Ultra-high-speed relays and fault current limiter operating

devices, where the signal-to-noise ratio is not well known.

d) Digital or low-energy protective relays, where signal-tonoise ratios are critical; performance evaluation of

different digital protective relaying algorithms is also

of interest.

e) Current and voltage transformers, which are critical to

the performance of high-speed protective relays.

f) Transient analysis programs, where improved fidelity of

computational results would be desirable. This could

include improved fault models of importance primarily

to ultra-high-speed relays and fault current limiter

control devices.

The laboratory system is applicable to the direct measurement of

extra-high voltage and current phenomena in a variety of areas. While

363
the present application will most assuredly advance the understanding of

utility power systems under transient conditions, future usage of the

laboratories and methodology can be easily transferred to other

disciplines where a knowledge of pulse power performance would be of

benefit.

7. Acknowledgements

The authors wish to express their gratitude for the technical

support provided by S. L. Nilsson of EPRI as well as all members of the

EPRI Advisory Committee on the Analysis of Transients on Transmission

Systems. W. H. Smith of Florida Power & Light should be recognized

for his work in the performance and direction of the design, implementation and installation functions of the project during his association

with Westinghouse. The authors also wish to acknowledge the valuable

efforts of Messrs. J. S. Barnick, J. E. Lokay, T. E. Mousseau, and

V. F. Wilreker of the Westinghouse Electric Corporation in analyzing

and improving the performance of the laboratories.

8. References

[1] IEEE Spectrum, "Technology '80: Power/Energy Developments,"

January, 1980, page 61.

[2] Shankle, D.F. and Smith, H.M., "Switching Surge Instrumentation

and Measurement Techniques for EHV Systems," (AIEE Winter General

Meeting, 1963, No. CP 63-75).

[3] Smith, W.H., Reed, T.J., et.al

.

, "New Field Laboratories for the

Measurement of Electrical Transients," (CIGRE Session of 1980,

No. 34-06).

364
APPLICATION PROBLEMS USING INSTRUMENTATION AMPLIFIERS

IN THE PULSE POWER ENVIRONMENT

E. Earl Cunningham

Ectron Corporation

San Diego, California 92111

1. Abstract

Difficult and unusual problems arise when low level signals

must be accurately measured in the pulse power environment.

Usually involved with signals immediately following the pulse,

this application demands unique capabilities of the signal

processing instrumentation. This paper discusses necessary

amplifier and signal conditioner characteristics in addition to

shielding and cabling recommendations. Test results involving

pulse common mode signals are presented to illustrate the

effects of improper shielding and cabling.

2. The Signal Problem

Much has been written about the application of

instrumentation amplifiers and signal conditioners in

conventional data acquisition systems [1,2,3]. However, signal

conditions present near pulse power systems are very unusual and

little material is available to aid the instrumentation

eng i neer. [5]

2.1 Desired Signals

Generally, when instrumentation amplifiers are used in this

application, they are amplifying analog data immediately

following the pulse. The desired signals are frequently

generated by more or less conventional transducers and, except

for the problems associated with the pulse, can be conditioned

and amplified using conventional techniques. Therefore, we are dealing with desired signal levels from microvolts up to several

hundred millivolts with a frequency range of interest of from DC

to 100kHz or so, and an accuracy of the amplified signal from a few percent down to 0.1% or somewhat better.

365
2.2 Noise Signals

Because of unavoidable coupling to the signal leads at the

source and into the cabling, plus common mode effects,

considerable pulse amplitude is usually present at the input to

the amplifier. Depending on a wide variety of factors the

amplitude of this pulse can vary from a few hundred up to many

thousands of volts with a duration from a few nanoseconds up to

several microseconds. At the amplifier this signal can be

either common mode (in common with the two signal leads), normal

mode (between the two leads) or, more likely, a combination of

both

Other noise signals likely to be present include:

Ring from the main pulse usually highly damped and much

higher in frequency than the desired data signal. This signal

is usually both CM and NM in nature.

A line frequency signal sometimes having high frequency

components that is predominately CM at the terminals of the

amplifier. Its amplitude can be millivolts to volts.

Pulse signals originating with switching devices ranging

from SCRs to switching power supplies. These signals, likely to

be both CM and NM, are especially troublesome if an A/D

converter is used following the amplifier.

Proper care in shielding and cabling such signals between

source and amplifier will minimize the effects of these noise

signals. However, if signal leads are improperly handled signal

and noise can be irretrievably mixed.

3. Shielding Considerations

Figure 1 shows an ideal situation where the source, the

amplifier and the signal leads are all enclosed within a single

continuous shield. This example illustrates a principle of

shielding that if all elements of a circuit are totally

contained within a single unipotential shield the existence of

other electrostatic potentials external to the shield cannot

influence the circuit within.

3.1 Faraday Cage

A term used to describe this type of shield is Faraday cage

or Faraday shield. In effect, such a shield transposes the

circuit from its remote location to the signal source and ground

point of the shield. A good shield is one that not only totally

encompasses the circuit within but also remains at the same

potential. This requirement demands that the primary guard

366
shield not be used to bypass drain currents and, of course,

requires that this shield be tied to only one ground point. If

significant current should flow in the shield it would no longer

be at signal potential and rejection of the common mode signal

could suffer.

FIGURE 1 IDEAL SHIELD

Note that if the amplifier of figure 1 had any connections

external to the shield the integrity of the shield could be

impaired. Since an amplifier usually has such connections (to

output instruments, to power line and to case) the amplifier

itself must maintain appropriate shielding between input and

these output connections. The degree to which the input must be

isolated is determined by the characteristics of the common mode

signal and the level and desired accuracy of the normal mode

si gnal

.

3.2 CM Voltages

DC Common mode voltages are usually easily handled in an input signal system. AC CM voltages are more of a problem since

shield capacitance to other ground potentials will cause shield

currents to flow. However, impulse CM signals are the most

difficult since even a small capacitance will cause large shield

currrents to flow. Therefore, as will be demonstrated later in

this discussion, shield impedance must be kept low both in the

cabling and within the conditioner in order to maintain proper

Faraday cage conditions.

367
4. Amplifier Characteristics

Most characteristics of available data amplifiers are

acceptable in the pulse power environment. These would include

linearity, dc drift, gain accuracy, noise, etc. Specifications

of the average amplifier that are not compatible with this

application would be common mode rejection at high frequency,

overload recovery both NM and CM, and NM and CM breakdown

voltage rating.

4.1 High Frequency CMR

High frequency common mode rejection is not only required

to minimize the effects of the pulse but also to reduce CM noise

that follows the pulse. To achieve high CMR at high frequencies

requires elaborate guard shielding of all input circuitry within

the amplifier. The degree to which all input circuit elements

must be guarded can be easily demonstrated by reference to

figure 2. C, represents the capacitance connecting one input to

case or ground at the amplifier. If a CM voltage, Ecm, is

present a current will flow through R

2

> the interconnecting

cable and C,. Because of the voltage drop across R~, the signal

from Ecm is no longer equal at the input terminals of the

amplifier and, of course, this differential signal is amplified.

A simple calculation will show that if Ecm has a frequency of

1MHz and R 9 is 100 ohms, C, must be less than 16 femtofarads

FIGURE 2 LEAKAGE CAPACITANCE

368
(10 farads) to provide a CMR of 100 db. Obviously, 100 db

of rejection is not enough to prevent CM overload during the

pulse considering the fact that high frequency components

present will far exceed 1MHz and the amplitude will likely be in

the thousands of volts. However, this amount of rejection will

minimize the duration of CM overload plus it will provide good

rejection of the smaller CM signals that follow the pulse.

Although difficult, guard shielding to this degree is possible

as illustrated by the Ectron Model 776. This unit's leakage

capacity, including its excitation supply, is approximately lOff

and 1 MHz CMR with 100 ohm unbalance is typically 109 db.

4.2 Maximum CM Voltage

The maximum common mode voltage rating of an amplifier

would normally involve a breakdown specification and, at a lesser voltage, an operating specification. Although it may not

be sufficient for many applications in the pulse power field,

practical constraints have limited the Ectron unit to ±3000

volts breakdown and ±1200 volts operating potentials.

4.3 Maximum NM Voltage

More difficult to achieve is a comparable normal mode

overload characteristic. Since a versatile instrumentation

amplifier must perform accurately with gains of 1000 or more an overload specification demands recovery down to the microvolt

level in a few microseconds. The 776 specification requires

that recovery from an NM overload of 1000 volts will be within

5% of final value in 10ys and within 0.1% in 30ys. Considerable

care must be exercised in selecting components exposed to a wide

voltage range not only within the amplifier but also in the

input signal system. To quickly recover from a 1000V potential

and accurately amplify microvolt level signals, dielectric

absorption and thermal emf's must be kept to an absolute

minimum. These precautions would apply, for instance, to

dividers or filters used ahead of the amplifier.

NM overload is also a problem in an excitation supply when

used in this environment. Protective circuitry must be

incorporated in the conditioner's design along with an

appropriate recovery capability.

5. Cabling and Shielding Recommendations

Because of the breakdown and recovery limitations of the

amplifier and signal conditioner, it is essential that pulse and

other noise signals be held to an absolute minimum if accurate

369
data is to be obtained. Much can be accomplished in reduction

of these noise signals by using proper cabling and shielding

techni ques

.

Generally, most of the pulse noise induced into the signal

leads will occur at the sensor and on the cabling leading to the

sensor since the radiation will be most intense at this point.

The sensor itself should have minimum cross sectional area and

should be oriented for minimum exposure. Preferably, the sensor

should be symmetrical such that the induced pulse will be CM

rather than NM.

Similar methods should be used in the selection and routing

of the wiring connecting to the sensor. The Los Alamos

Scientific Laboratory has done considerable work in developing

flat cable for this use, primarily for applications at the

Nevada Test Site. When oriented properly, these cables expose a very small cross sectional area to the beam. Again, cable

symmetry and routing will reduce NM pickup.

Shielding of the sensor and leads connecting to the sensor

is extremely important in order to keep pulse pickup down to

tolerable levels. Although frequently limited by constraints of

the experiment, liberal use of a shield material effective over

the spectrum involved is highly recommended.

Tables 1 and 2 give general cabling and shielding

recommendations for signal wiring beyond the sensor area. These

suggestions follow the rules of standard practice for

instrumentation systems but with modifications to fit the pulse

power environment.

Cable selection is of considerable importance in a typical

system not only because of the effect on signal quality, but

also because of the significant cost involved when long cable

runs and numerous channels are required. Too, this application

often demands the highest quality cable.

Note that double shielding of each signal cable is

recommended with even triple shields advantageous. Each shield

reduces drain problems and offers additional grounding choices.

Low shield inductance and resistance is important in being able

to drive guard areas in the amplifier/conditioner at pulse

frequencies. Foil type shields provide total shield coverage,

however, they usually require a separate drain wire because of

higher resistance. Use of twisted pair cables having good

symmetry is important to minimze the effects of magnetic fields.

This characteristic also contributes to making the capacitance

of each lead to the shield equal which is necessary to prevent

CM to NM conversion within the cable.

370
TABLE 1. Cabling Recommendations

Use twisted pair double shielded cable.

Keep shield inductance and resistance to a minimum.

Capacitance between each conductor and shield should be

equal [5].

Do not coil excess cable.

Minimize capacitance between cables and ground surfaces.

Do not use steel core cable.

Voltage rating of the cable must be compatible with the

appl i cat i on

Avoid areas of high magnetic fields when routing cable.

TABLE 2. Shielding Recommendations

Maintain total shield integrity from source to amplifier.

Connect shield to zero signal point having low source

impedance. "Ground" shield at only one point.

Minimize shield current. Use drain wire or second shield.

Each shield should contain only one signal system.

Bundle several cables carrying similar NM and CM signals and

cover with overall low inductance sheath.

Use Faraday cage configuration if possible.

6. CM Pulse Tests

The following tests, using an Ectron Model 776B

Conditioner/Amplifier, will serve to illustrate the effects of

less than ideal shielding and cabling conditions.

Figure 3.1 shows the basic test configuration. A bridge

transducer was simulated using four 120ft resistors. Two

Trompeter Quadrax cables, each 25 meters long, interconnected

the excitation supply and amplifier to the bridge. The

excitation was set to 10 volts and each of the test

configurations was tested at amplifier gains of 20 and 1000.

The pulse generator, simulating a CM signal, produced a 730ns

pulse (FWHM) having an amplitude of 62 volts with rise and fall

times of under Ins. No attempt was made to terminate the cable.

Figure 3.2 is an oscilloscope trace showing amplifier

output with a sensitivity of 1 volt per division, sweep speed of

5ys per division and an amplifier gain of 1000. These

connections, following recommended grounding and shielding

procedures, produced an output perturbation of only 0.5 volts

settling to 1% in 4ys. (Additional scope pictures were taken at

higher sensitivities allowing 1% measurements to be accurately

made.) Resulting CMR is 102db. Note that increasing the

amplitude of the CM pulse to 1000 volts will increase the

371
resul t i ng

overl oad

730ns used

occ ur

output to approximately 8 volts, still well under

Furthermore, if pulse width is decreased from the

here, a similar decrease in output amplitude will

2 5M

FIGURE 3.1 TEST CIRCUIT 1

«,J

1

1

\k

r

1

FIGURE 3.2

To illustrate the need to maintain a low impedance

connection to the guard of the amplifier, a 10ft low inductance

resistor was added between cable shields and conditioner guard.

This resistor was carefully mounted so that shielding was still

complete. Figure 4.1 shows the circuit and figure 4.2 shows the

resulting trace. Note that amplitude increased to over 2 volts

and recovery to 1% increased to 8ys. These increases are caused

by the amplifier's guard shield potential lagging the CM

potential of the signal leads causing current to flow in the

372
signal leads. This current produces a NM signal which is

amplified by the amplifier.

The need for total guard integrity is illustrated in

figures 5.1, 5.2 and 5.3. Here, a 5 picofarad capacitor was

connected between one input lead at the amplifier and output

ground. This capacitance is similar to running signal leads

through a connector whose shell is not connected to guard. The

lower sensitivity trace of figure 5.3 shows a maximum amplitude

excursion of 5 volts. One percent settling time is 8ys.

373
FIGURE 5.1 TEST CIRCUIT 3

FIGURE 5.2 FIGURE 5.3

A fourth test was performed with the guard tied to output

ground keeping all other connections as shown in the first test.

This arrangment , of course, voids the protective guard shielding

within the amplfier, and substantially reduces amplifier CMR.

Because of the effects of saturation and considerable ringing,

settling time to 1% increased to 22ys. Indicated CMR decreased

to 75db although this figure should be less since saturation

limited the true output excursion. When this test was repeated

at a gain of 20, recovery to 1% improved to 13ys but CMR was

only 46db. Again, if pulse amplitude was increased recovery

time would proportionally increase and, similarly, if the pulse

was narrowed recovery would improve.

374
These four tests, summarized in table 3, illustrate the

need to carefully consider the shielding and cabling of signal

conditioning equipment when impulsive CM signals are involved.

Although each of these tests involved only a single cabling

deficiency to best illustrate the results, an actual test

configuration can be a combination of these problems with

usually an additive effect on recovery and CMR.

TABLE 3. Test Summary

Test 1 Shield

to Guard

Test 2

10ft Shield

to Guard

Test 3

5pf Cap.

Test 4

Guard/Ground

Gain X1000 X1000 X1000 X1000

Max

.

Overshoot .5V 2.2V .5V 11V (sat)

Time to 1% 4yS 8ps 8yS 22ys

CMR 102 89 82 75

7. Faraday Cage System

The Faraday cage concept has been applied to the use of

instrumentation amplifiers in the pulse power environment.

Although not easy to accomplish, this shielding approach offers

significant advantages over more conventional cabling and

shielding methods.

Figure 6 shows a typical data channel connected in this

manner. The instrumentation is all contained in an isolated

enclosure which is cabled to the signal source using proper

shielding procedures. Thus, the source, the signal leads and

all of the instrumentation are totally encompassed within a single shield system that is strapped together using very low

impedance drain connections. As shown, the only connections to

ground are the stray capacitances which would normally be

located mostly at the source and at the instrument enclosure.

Power must be coupled into the enclosure using an isolation

transformer and the signal transferred out using suitable

isolating techniques such as fiber optics or properly designed

signal transformers. The improvement that the Faraday cage

offers over a conventional system is in direct relation to the

ratio of drain impedance to the impedance of the capacitance of

the instrument enclosure. Therefore, it is important to keep

the capacitance of this enclosure with respect to ground to an absolute minimum. The use of the Faraday cage system has proven

successful in tests both at the Nevada Test Site and at Sandia

on the REBA Marx generator. Where the situation permits, this

cabling concept is highly recommended.

375
LOW IMPEDANCE SHEATH CONTAINING SEVEPA'U CABLES

GROUND AT INSTPOM£NTATI ON SITE FIGURE 6 FARADAY CAGE SYSTEM

8. Conclusions

The amplification of low level signals in the pulse power

environment requires special care in cabling and shielding,

particularly at the source, to minimize pulse coupling onto the

signal leads. The amplifier and signal conditioner used must

have unique characteristics to perform well in this environment.

Where conditions are particularly poor or where highest

performance is required, the use of the Faraday cage is

recommended. This approach encloses source, cabling and

instrumentation within a single isolated shield system and uses

a fiber optic link to couple out the amplified signal.

376
9. Acknowledgements

R. S. Dingus of Los Alamos Scientific Laboratory

contributed information on the use of flat cables and sensor and

cable shielding. Raymond M. McNairy of Lawrence Livermore

National Laboratory supplied data regarding operation of the

Faraday cage system.

10. References

[1] Morrison, R., "Grounding and Shielding Techniques in

Instrumentation, Wiley and Sons, New York

[2] Harper, C.A., "Handbook of Wiring, Cabling and

Interconnecting for Electronics", McGraw-Hill

[3] Engle, C.E., "Techniques to Analyse and Optimize Noise

Rejection Ratios of Low Level Differential Data

Systems", Dana Laboratories, Inc., Irvine, CA.

[4] Crawford, T. , McNairy, R., "High Performance Signal

Conditioning Unit", University of California

Specification No. J8-76-173, June, 1977

[5] Cunningham, K.E., "Effects of Impulse Common Mode

Signals on High Performance Amplifiers", Final Report

for LASL Contract No. 4-L60-5011M-1 , Ectron Corp.,

Feb. 1980

[6] Trom peter, E., "Electronic Systems Wiring and

Cabling", Pgs 1-7 of Catalog T-13, Trompeter

Electronics, Chatsworth,

Ca.

377
Discussion Session 7

Bob Hebner (NBS). Question to Terry McComb. You and I have performed

very similar measurements of waveforms and attempted to reconstruct

waveforms. I interpreted what you were saying at the end of your

presentation to mean that you felt that the residual disagreement

between your calculated and measured outputs was due to a problem in

the calculations and that you mentioned that you were going to have

to work harder on the calculations. I concluded that our problem was

not in the calculations, but in the measurement of the step response.

If you give me freedom to choose my step response, I can always make

the calculations agree with the measurements. So, my question then

is do you indeed feel that your problem is in the calculation rather

than in the measurement?

Terry McComb (NRC, Canada). Well, I think that they are really tied

together. 0~ne problem was that the wave was relatively long compared

to the significant part of the step response. So the step response

was changing significantly only for the first part of the sweep.

From that point of view then, we are doing the correction in one of

the more disadvantageous regions. When we move to faster impulses

and we also record the step response on the same scale, then we will

have significant information right through the sweep. So our corrections will be bigger, but we would expect them to be more

accurate. Another problem was that in order to generate the impulse

response, one has to differentiate the step response and this introduces

more noise.

Richard Malewski (IREQ). I think that one problem comes from the

fact that our A-to-D converter has a finite sampling time. The

samples were taken at 10 nanosecond intervals which obviously implies some filtering of the signals. That was one possible source of

inaccuracy in reconstruction. We believe that with the new model

which now is becoming available with 5 ns sample times, we will be

able to produce more exact, more accurate reconsti tution of the

measured impulses.

Bob Kolibas (Raytheon). Mr. Cunningham, I enjoyed your presentation.

On one of your last slides showing the Faraday cage I didn't notice a return to the shield. This implies that it was floating from any

return or ground and I am confused. Was it truly grounded some place

to drain off the currents?

Earl Cunningham (Ectron Corporation). Maybe I should look carefully

at the drawing, but indeed the whole Faraday cage must be isolated

from ground at the instrumentation site. The slide in question is

figure 6. It is true that the inner shield of the signal cable

terminates at the guard of the amplifier and therefore has no return.

However, the amplifier can withstand a very high pulse voltage between

guard and case or output, plus we have very heavy cabling interconnecting

the case that encloses the instrumentation and the low impedance driving

378
point at the source. Thus, the common mode signal present on both

signal leads and shield at the amplifier will also be applied to the

instrumentation case through the heavy cabling and bonding. If this case has low leakage capacitance to ground than the cable driving it

will be able to minimize the common mode signal that the amplifier

must endure. In other words, if the ratio of capacitive reactance to

cable impedance is low then the perturbation between signal input and

output common of the amplifier will be low which will minimize the

task the amplifier has to perform to recover quickly and to produce

accurate data.

Malewski . We once studied this problem. We have two boxes and want

to send a signal via coaxial cable from box A to box B. We know that

there is a common mode interference signal applied to the two boxes

that the current circulating in the shield of the cable produces an error. The way of thinking is very much as presented here and is to

put in parallel a low impedance conductor which splits this current

into two parts. The large current we will call Ig and small current

I c

. The efficiency n of this system can be expressed with the ratio of

I c over Ig. We looked for the optimal geometry of this ground cable

in order to bring this ratio to the low value. The typical

configuration was a pipe representing coaxial cable shield and a flat

copper shield. We examined two parameters, the distance between

the cable and the ground h and the width of the copper strap.

As a function of frequency, our results showed that the closer the

cable is to the copper strap, the better the system, the lower the

n parameter. So if we take a different value of h and reduce the

width of the copper strap then again the n will increase. The flat

bar is more efficient than the round conductor, i.e., for the same amount of copper, we can get much better performance with the flat

rather than the round conductor. There is a publication on this

subject giving the specific values for the experiment performed.'

Jack Katzenstein (Maxwell Laboratories). I would like Mr. Poliner to

comment on the role of the electro-optic sensors in the utility

instrumentation room.

Randy Pol iner (Westinghouse) . This is a pretty general topic. We

see it as the future for the utility environments just because of

the isolation and data integrity. We are demanding more and more from

our data acquisition systems; particularly advanced relaying schemes

have made it almost a necessity to have the isolation gained from

electro-optics.

1 R. Malewski , G. Nourse, "Transient Measurement Techniques in EHV

Systems." IEEE Transactions, Vol. PAS-97, No. 3, May/June 1978,

p. 893-902.

379
John Anderson (GE). I just had a couple of very short comments. I am wondering whatever happened to all of the ferrite cores that people

used to put on cables. It seems to me that this really helps. The

second comment is directed to Mr. Poliner. We have used a cascade of

ferrite core transformers, to bring power from the ground up to the

signal electronics. Working at 30 kilohertz we can get efficiency

along the transformer cascade in the order of 80 percent and overall

efficiencies of the system of about 40-50 percent from the power line

at ground to the electronics at the top. This has been very effective

as far as we are concerned. I would like to introduce this as a possible approach for people who want to get power to their electronics

in a very very high voltage environment.

Poliner . There are several schemes to get power to floating electronics.

The ultrasonic rod we used is, of course, just one of them. People

have used wind turbines, solar energy, and some have gone so far as to use broom handles and motor generator sets which work very nicely

for some applications. Our particular application required that we

be totally isolated from 60 hertz interference so that any pulses

that we were trying to measure would not be in turn affecting our power supply. That is one of the main reasons that we did have to go

to the ultrasonic rod technique, although admittedly it is a very

poor means of transferring power in terms of efficiency.

380
DIGITAL CORRECTION OF CABLE ATTENUATION LOSSES*

C. Trivelpiece, R. Richardson, J. Shannon, J. B. Smith

Maxwell Laboratories, Inc.

8835 Balboa Avenue

San Diego, CA 92123

The use of sensitive electronics on large pulsed power generators

often necessitates long (>50 ft) signal cables. Many of the signals

measured have risetimes less than 10 ns. The long cables cause a significant degradation in the risetime and shape of these signals.

Attenuating low freguencies using a passive compensation network is one solution to this cable loss problem. However, when data acquisition is accomplished by computers and high speed transient digitizers,

it is possible to accurately correct for cable attenuation without the

problems associated with passive compensators. We will discuss a technigue for using Fast Fourier Transforms and waveform correction

without introducing additional errors.

*Work supported by the Defense Nuclear Agency.

1. Introduction

Because of the large amounts of EMI generated t>y high voltage,

high power machines, it is necessary to put the data recording devices

such as oscilloscopes and transient digitizers in a screen room. This requires cable runs of 50-100 ft. In addition, there are sometimes short runs of lower quality cable in the screen room to allow

for trigger delays, fiducial marks, etc. These cable runs lead to

degradation of the high frequencies which lead to risetime and/or

peak amplitude degradation.

In the past, passive compensation networks were used to correct

this problem. At present, the trend for those who can afford it, is

to use fast transient recorders and on-line computers to acquire the

data= In this situation, many of the frequency compensations, including the cable response, can be handled in the computer software.

This takes less time to achieve the compensation and cable changes

can be accomodated by software changes rather than a physical circuit.

381
In this paper we describe a method used at Maxwell to correct for

cable losses. We will present an analysis justifying the method and

present experimental data showing the accuracy of the correction.

2. Analysis

In this section the mathematical model used in the analysis will

be described. The attenuation in coaxial cables has been treated

analytically before [l] . However, it is necessary to develop it here

in some detail to illuminate the method used in the correction.

It is well known, that the attenuation (in DB) in coaxial cables

is given at sufficiently high frequencies by

DB ~ Vf~ , (1)

where f is the frequency. This is a consequence of the copper loss

in the cable, since at sufficiently high frequencies the resistance is

proportional to y/f . In Laplace transform space, the transmission

line equations can be written as

dV

dx

dl_

dx

= L/I + a y/sl

(2)

= C/sV

where V and I are the transformed voltage and current, L" , C

are the inductance and capacitance per unit length, a is a constant

of the cable describing the resistive loss at a "frequency" s . The

constant a is given by

a = j- J-4 (3)

where l±is the total length transverse to the direction of propagation, a is the conductivity of the cable and y

Q

is the permeability

of free space. 1

Equation 2 has solutions of the form

(4)

332
where y is the propagation constant given by

c 2 3/2

where u is the speed of propagation in the cable and Z is the

impedance of the cable given by 0

_ < 6 )

z o Nr

For sufficiently high frequencies, y can be expanded to give,

1/2 2

Y

u 22 8 - 2 ••• l/j

0 Zo

For any practical cable and reasonable frequency, all terms except the

first two can be neglected.

In this case, a wave propagating down the line has the form

V, I exp

sx

u as

1/2

2 Z, (8)

The first term in the argument is the result of the delay time

along the cable and the second term gives rise to the dispersion.

In actual fact, one must be a bit more careful here because of

the input and output impedance at the ends of the cable. Because of

the frequency dependent series resistance the characteristic impedance

defined by the motion of the voltage and current wave is not Z Q

but

383
This means that reflections can occur from a line terminated in Z In most cases of interest at Maxwell, the line is fed from sensors

0

that effectively have Z0 as an internal impedance and are likewise

terminated in Z 0

. For this case it can be shown that the output

voltage is related to the sensor voltage by

v

= y _Ayus_

out 9 (yu + s)

2 -yx

(10)

where V q is the sensor voltage normalized to give Vout on a dispersion-less line. If the same expansion of y is used

out

~ s

( 1 + au • 2 ZpNfs )

I 2Z

0

Vs )

exp

sx

u axs2 Z.

1/2

]

(11)

exp

y

sx

u axs2 Z

1/2

o J

Hence the dispersion can be described using only the exponential term.

3. Waveform Correction

As was seen above, the voltage at the end of a line of length x is related to the input by

(- -O V out

V

in exP'- I < 12 >

334
_ c y where the delay term e /u

forward manner we can solve for

has been suppressed. In a straightin

= V

out

exp

faxs 1/2

2Z. (13)

Formally, one would like to transform this to the time domain and

convolve V ou t(t) with the inverse of the exponential term and get

V-j n (t).. Unfortunately, exp (axs%/2 Z

p

) is not mathematically suitable for transforming to the time domain although its product with

out

is transformable.

If only modest corrections to the waveform are required (<20 percent or so) there is an approximate way of finding V-j n (t) . A small

correction in the time domain means that in the frequency domain the

argument of the exponential is small. We can then write

Vv . = V 11+

in

v out I1 2 Z i1 + O (14)

This can be written,

V. = V + sV ± in out \ out (15)

We notice that multiplying by s is the same as taking the derivative

in the time domain and that the inverse Laplace transform gives

SB

-1/2 (16)

in the time domain. Equation 15 then transforms to

385
dV

out

0

where

K(t) = ttt- t^t t > o

2 Z o v^T

(18)

= 0 t < 0 .

Equation 17 is a straight forward operation with the software of

the most modest computer. It will be seen below that the approximations made leading to Eq. 17 are reasonable and this correction procedure works remarkably well.

4. Explanation of Computer Code

The waveform data from the digitizers is a series of 512 points.

A mathematical model is a series of delta functions separated by a constant time interval At . Also, the vertical resolution of the

digitizers is at most 512 points but in practice probably around

200-400 points. This tends to cause a staircase representation of the

signal. When the derivative is taken a lot of spikes and apparent

noise are generated. A three point average is done on the waveform

before the derivative is taken to cut down on the noise produced. The

three point average was done by averaging each point with the points

on either side of it. If the end points of an array are not zero they

will produce large spikes when the derivative is taken. This can be

eliminated on the derivative waveform by making the last three end

points equal to the fourth point in from the end.

The correction table is generated and the convolution integral

performed. The correction is then added to the original waveform.

10 REM THIS CODE WILL PERFORM A CABLE CORRECTION

20 WAVEFORM II IS 1(511), DI,HI$,VI$

30 WAVEFORM VV IS V(511) ,DV,HVS,VV$

40 WAVEFORM 00 IS 0(511) , DO, H0$ ,V0$

50 WAVEFORM GG IS G(1023) ,DG,HG$,VG$

60 DIM IP(511)

70 REM II IS THE INPUT WAVEFORM

80 IP=I

90 REM DO A 3 POINT AVERAGE TO SMOOTH WAVEFORM

100 I(l:510)=(I(0:509)+I(2:511)+I(l:510))/3

336
110 REM TAKE DERIVATIVE OF INPUT

120 DIFF II, VV

130 REM GET RTO OF END POINT PROBLEMS FROM DERIVATIVE

140 V(0:2)=V(3)

150 V(509:511)=V(508)

160 REM GENERATE CORRECTION TABLE

170 FOR J=l TO 511

180 T=J*DI

190 0(J)=2.3E-07*(126/SQR(T))

200 NEXT J 210 0(0)=0

220 REM DO CONVOLUTION INTEGRAL

230 CONVL 00,VV,GG

240 REM ADD CORRECTION TERM TO ORIGINAL WAVEFORM

250 V=IP+G(0:511)

5. Implementation of Computer Code

To generate the correction table a few physical factors about the

cable must be found. The length can be easily measured and the imped- ance can be looked up in a book. Now a constant for the attenuation

of the cable is needed. Although the correct number for the attenuation must be found experimentally, an approximate number can be found

using the attenuation versus frequency curves published for cables.

This calculated factor agrees closely with the empirically determined

one and provides a useful starting point for the experimental determination.

To find the approximate attenuation factor for the corrections

the slope in ln( In/output) versus \/w (w = 2irf) must be found from

the db loss curve. Since the slope is a straight line in the frequency range of the tables only two points are necessary. Example:

RG-58 has a 15 db/100 ft loss at 540 MHz and a 3 db/100 ft loss at

35 MHz. Transforming these into the natural log of the attenuation

and Vw~ gives

15 db 1.72 0-0172

0 54Q m 5 Q2 1Q4 (ig) 100 ft 100 ft ft 1 D4U 0 iU u ;

3 db 0.345 0.00345 @ 35 MHz - 1.48 x 10

4

(20)

100 ft ~^100 ft

~* ft

eiTM 1.72 x 10" 2

- 3.45 x 10" 3

, 3.16 x 10" 7 slope = ^ ^ - ^

(21)

5.82 x HT - 1.48 x 10

4 TL

387
Attenuation

2 slope _ 2.52 x 10

7T X ft ft

-7

constant = (22)

the correction table has the form

K(t) = 2.52 x 10" 7 x Len9 th (ft ) (23)

. / T

The value used in this paper for a length of 126 ft of RG-58 was 2.3 x 10" 7 . A value for RG-213 was computed to be 9.49 x 10~ 8 . The

factor for the loss in two different types of cables that are hooked

together is approximately the sum of their lengths times their respective attenuation constants. The correction table for 126 ft. of RG-58

is shown in Figure 1.

3

2.5

2

0.5

1

0 0 10 20 30 40 50 60 70 80 90 100 1E-9 SECONDS

Figure 1. Correction table for 126' of RG-58.

The test setup used is illustrated in Figure 2.

388
PULSE GENERATOR

HIGH SPEED DIGITIZER

1 Mn INPUT

BNC

i 'TEE" /

50 n

TERM

1 0,00.00/ TEST CABLE

Figure 2. Test set-up for measuring cable response.

The vertical amplifier has a 1 mQ input impedance and the

"Tee" is a regular BNC "Tee". This way the input and output of

the cable can be measured without affecting the circuit.

A series of pulses are recorded and the attenuation factor is

changed until the best fit is obtained. For best results the test

should be done at the same sweep speed that will be used to collect

data.

6. Results

In Figure 3 a fast rise square wave was put into a 126 ft cable

of RG-58 and the output was measured (dotted waveform). The correction was applied to the output of the cable and the waveform with

squares was produced.

Risetime Peak Value Integral

Input 1.82 ns 100% 100%

Output 12.6 ns 82.5% 82.8%

Corrected 1.89 ns 99.3% 98.2%

389
12

10

INPUT ^ OUTPUT A CORRECTION

j ,

. —

/'

/ j\

1 \\

/

'\ V

\\ *v

SQUARE WAVE INTO 126'

i i i i i i

RG-58

i i i 0 5 10 15 20 25 30 35 40 45 50 1E-9 s

INPUT OUTPUT

CORRECTED

R I SET I ME

1.82 ns 12.60 ns 1.89 ns

PEA K VALUE

100.0%

82.5%

99.3%

INTEGRAL

100,0%

82.8%

98.2%

Figure 3. Cable correction for fast square wave

Figure 4 shows the correction that was added to the output of the

cable to simulate the input.

0 5 10 15 20 25 30 35 40 45 50 1E-9 SECONDS

Figure 4. Correction that was added to output in Figure 3.

390
Figure 5 shows a more gentle pulse passing through the cable.

1.1

1.2

1,0

0,8

V

0,6

0,4

0,2

0

0 5 10 15 20 25 30 35 40 45 50

1E-9 s

Figure 5. Pulse into 112' RG-58.

7. Summary

For data collection systems that are using high speed digitizers

and digital computers, this method provides a convenient and accurate

way of compensating for cable losses and distortions. This correction

is accurate for a wide range of cable lengths and pulse shapes.

8. Reference

[1] R. L. Wilmington and N. S. Nahman, "Transient Analysis of

Coaxial Cables Considering Skin Effect", Proc. IRE pp 166-174;

February, 1957.

INPUT OUTPUT

CORRECTION

391
PRECISION PICOSECOND-MICROSECOND ELECTROMAGNETIC

WAVEFORM MEASUREMENTS AT NBS

R. A. Lawton

Electromagnetic Technology Dvision

National Bureau of Standards

Boulder, Colorado 80303

Continuing research in the measurement science of electromagnetic

waveforms has resulted in the development of an Automated Pulse Measurement System (APMS) . This system consists of a sampling oscilloscope

mated with a minicomputer to provide for automatic acquisition and

processing of repetitive pulses with durations and transition times in

the picosecond to microsecond time frame. In addition, work has been

started to add single transient capability to the APMS.

The development of this capability has required the careful development of the electronic circuit interfacing between the sampling oscilloscope and the minicomputer and the development of the appropriate soft- ware to allow the scope to be properly controlled by the computer.

Vertical and horizontal calibration techniques have been developed

which provide an order of magnitude improvement in accuracy.

Reference waveform generators have been developed and new ones are contemplated to serve as a check on the complete measurement system.

Finally, a measurement system analysis has been performed which

includes system modeling, signal parameter characterization and optimal

data processing algorithm identification.

With the APMS we are now able to characterize noise free, repetitive

time domain waveforms with an uncertainty of 1% of full scale in amplitude and 0.5% of full scale on the time axis.

In addition, with our best waveform generators, we can do characterization in the frequency domain over the frequency range 5 MHz to 10 GHz.

392
PRECISION PICOSECOND-MICROSECOND ELECTROMAGNETIC

WAVEFORM MEASUREMENTS AT NBS

R. A. Lawton

Electromagnetic Technology Dvision

National Bureau of Standards

Boulder, Colorado 80303

Since time resolved waveform measurements are currently assuming

increased importance in the electromagnetic community, we at the National

Bureau of Standards (NBS) have an on-going measurement program to develop the tools needed to put waveform measurements on a firm foundation.

Much of the development work at NBS has been concentrated on repetitive pulses and low energies [1,2], but the needs for high energy single

transient measurements have not been forgotten [3,4].

The development of modern waveform measurement instruments such as sampling oscilloscopes and transient digitizers have greatly aided the

waveform development program at NBS, however, the development of modern,

dedicated, interactive computers for data acquisition and processing has

been of equal importance. These computers have made possible the acquisition of 1024, 2048 and 4096 etc. point waveforms with comparative ease

and speed - in a fraction of a second. This acquisition time is so quick

that the same waveform may be reacquired hundreds of times to allow for

signal averaging all in just a few seconds.

I will now describe some of the measurement systems developed at

NBS for the precision measurement of time domain waveforms. This will

include a description of the waveform acquisition equipment and its

modeling, data processing and signal analysis.

The most highly developed waveform measurement system at NBS is the

Automatic Pulse Measurement System (APMS) which consists of a sampling

oscilloscope controlled by a minicomputer for making measurements on re- petitive pulses. The block diagram of this system is shown in figure 1

(PTR denotes paper tape reader and PTP denotes paper tape punch) and a photograph of the system in operation is shown in figure 2. The A/D

converter digitizes the amplitude of each sample which is then stored in

memory after which the computer through the D/A commands the oscilloscope

to take another sample. This system can accurately record waveforms which

have transition times as short as 20 picoseconds. The measurement precision is determined by the amplitude noise and timing jitter of the

system. This system has been characterized for these parameters and

found to have standard deviations of 2 millivolts of amplitude noise and

6 picoseconds of timing jitter for 10 mV/cm vertical sensitivity and

200 ps/cm horizontal sensitivity, using a 200 MHz precision oscillator

as a clock. The APMS is capable of measuring pulses with amplitudes in

the neighborhood of 1 to 500 mV. Higher amplitude signals may be measured

at reduced accuracy using calibrated attenuators.

393
394

Once the data is acquired and stored, data processing is available

in the form of the fast Fourier transform (FFT) to obtain frequency

domain information, deconvolution to remove distortion introduced by the

measuring instrument, and signal averaging to extract low amplitude

signals from noise.

Examples of these types of operations will now be given. Figure 3

illustrates the acquired waveform of an NBS pulse generator and its

spectrum obtained by the FFT. Figure 4 shows the impulse and step re- sponse of a 65 nanosecond coaxial delay line used with pulse generators

where no pretrigger pulse is available. Figure 5 shows the waveform of a pulse fed through this delay line before and after the distortion due

to the line has been removed by deconvolution. Finally, figure 6 shows

a waveform that has been attenuated below the noise level (upper trace)

,

and the result of signal averaging of many repetitions of the same waveform (lower trace).

Now how does this capability apply to high energy transient measurements with waveforms that are frequently of a non-repeating character?

The answer lies first in the ability to accurately determine impulse

responses with the APMS as was done for the delay line in the example

just given. More appropriate to high energy measurements, the impulse

response of attenuators can be obtained and then used to deconvolve the

distortion of the attenuator from the transient measurement. Secondly, we have made an initial approach to developing standards for single

transient measurements. This has consisted of acquiring a low light

level vidicon which among other things can be used to digitize the trace

of a fast, real time oscilloscope as shown in figure 7. The acquired

waveform can then be stored on a disk and is available for future processing in the same fashion as the repetitively pulsed signals described

earlier (except for signal averaging) or for display on a monitor like

the one shown in figure 8.

We have also done some signal processing analysis of a commercial

transient digitizer in connection with our development of a laser pulse

power meter. This power meter has been described elsewhere [3], but

consists of a box housing an optical diffuser type attenuator and PIN

photodiode detector shown in figure 9 and a transient digitizer shown

in figure 10. This description includes the determination of the system

impulse response, attenuation factor and a routine for calibrating the

transient digitizer response for both amplitude and time base.

Data processing that has been used at NBS with the transient digitizer includes numerical integration in calculating pulse energy in the

attenuator measurement, automatic calibration correction, and computed

input waveform in terms of power versus time.

To improve the accuracy of both repetitive and single transient

measurements we have developed an automated calibrator for pulse amplitude and time scales [5], standard waveform generators [6] to test pulse

measurement systems, system modeling and deconvolution techniques [7],

signal parameter characterization techniques, and optimal data processing

396
Figure 3. Acquired Waveform (top) and Spectrum (bottom) of Pulse

Generator.

397
rn

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393
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399
CD

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400

402


algorithm selection techniques [8].

It is not my purpose here to describe these techniques in detail.

I will leave that for those who look up the references. However, I will

make a few comments about certain of these techniques.

By using the automatic calibrator the uncertainty in amplitude has

been reduced to < 1% of full scale and the uncertainty in time has been

reduced to < .5% of full scale. The use of reference waveform generators

which have a smooth gaussian transition from the 0 to 100% level without

any overshoot or ringing make it possible now to characterize other waveforms over the entire observation time window instead of only specifying

the initial 10% to 90% transition time (rise time). In addition the

specification of the 10% and 90% points is unambiguous for this type of

waveform, an example of which is shown in figure 11. By applying system

modeling and deconvolution techniques we have been able to extend the

range over which we can characterize pulse generators in spectrum amplitude from a previous maximum frequency of 3 GHz to 10 GHz at present.

The goal of all of our measurement activity is to be able to provide

for you, the customer, measurement services. These most frequently take

the form of calibrations. At present NBS offers calibrations in transition duration (risetime) of both pulse generators and low-pass filters,

pulse delay, spectrum amplitude, and wideband attenuation.

Various research efforts are underway to improve our waveform measurement and calibration services. Present research efforts include

further refinement of sampler modeling and real time data acquisition

research to select optimum filtering algorithms for the deconvolution

process, research in jitter characterization and new sampling technique

development using Josephson Junction switches and reference waveform

development using photoconductive switches and other solid state devices.

Using all of the above developments and resources at the disposal of

NBS and that of our colleagues, waveform measurements are beginning to

reach the status of a measurement science with a firm foundation.

405
auo

14-1 a)>

a)

0)oC

<uM

QJ IW

QJ

•HpHI

0} •H

cn03

ca o

43

4Joo6

5-1 3

•H

406
REFERENCES

Gans, William L. , "Present Capabilities of the NBS Automatic Pulse

Measurement System," IEEE Trans. Instrum. Meas., IM-25, pp. 384-388,

(Dec. 1976).

Andrews, James R. , "Automatic Network Measurements in the Time

Domain,", Proc, IEEE, Vol. 66, pp. 414-423, (April 1978).

Young, M. , and R. A. Lawton, "Measurement of Pulsed-Laser Power,"

NBS-TN-1010, (Feb. 1979).

Nahman, Norris S., "Picosecond-Domain Waveform Measurements,"

Proc. IEEE, Vol. 66, pp. 441-454, (April 1978).

Andrews, J. R. , and E. E. Baldwin, "Amplitude Calibrator for

Oscilloscopes," To be published.

Andrews, J. R. , "Pulse Reference Waveform Standards Development at

NBS," (Invited), Proc. ATE Seminar /Exhibit , Pasadena, CA, pp. IV13-19, (Jan. 1981).

Nahman, N. S., and M. E. Guillaume, "Instrument Modeling and

Deconvolution Improve Time Domain Calibrations," NBS Dimensions.

To be published.

Nahman, N. S., and M. E. Guillaume, "Deconvolution of Duration

Limited Waveforms," Digest of Papers, URSI/U.S. National Radio

Science Meeting, Boulder, CO., (Jan. 1981).

407
Discussion Session 8

Bob Hebner (NBS). I have a left-over comment and question about this

morning's paper. The comment that I wish to make first was on Dr. Malewski's review of his paper in which he had studied the magnetic

coupling effects over a thin plate as opposed to a wire. He came to

the conclusion that a strap was a better way of grounding when you

are trying to get good signal characteristics. I think that he owes

it to Mr. Poliner to at least acknowledge that that was exactly the

point that Mr. Poliner had made earlier in his presentation. Then I wish to turn around and ask a question of Mr. Poliner. He mentioned

in his presentation that the limiting accuracy that he had now was

in the A-to-D converters. I assume that means the limiting accuracy

you have in the low voltage part of measurement system. It seems to me that at least in the voltage channels the accuracy limitations

would be imposed by the divider that you are using to get from high

voltage to low voltage.

Randy Polin er (Westi nghouse) . I appreciate your pointing out the

fact about copper strapping. I would like to ackowledge Bob Hebner's

aid in describing that phenomena as a member of our Electric Power

Research Institute (EPRI) review committee. As far as the absolute

accuracy of the system, you are correct in assuming that the A to D

accuracy mentioned is in the low voltage section. In the case of the

voltage transducers, however, the primary transducer and low voltage

section accuracies are about equal.

Neville Harris (Ion Physics). Just a comment with regard to the

correction of cable losses. My experience with cables has been a bit

sadder. I haven't so much worried about attenuation, but what has

concerned me are reflections within the cables. Eccentricities are sometimes introduced in the manufacturing process and the impedance

variation of cables along their length can be very significant,

particularly in the very low gigahertz region. The impedance variation

can be either random or periodic in nature and I have had many sad

experiences with cables. I was wondering whether you tested each

cable separately and didn't just correct the loss but also threw out

bad cable?

Craig Trivel piece (Maxwell Laboratories). Well, you know I tested a series of cables. I didn't run across any of those problems. Perhaps

we had a slightly lower frequency range. Most of our pulses have

risetimes of 10 to 50 nanoseconds so I guess that it just didn't show

up.

Bill Boyer (Sandia). We do test all of our cable with a TDR and we don't use the bad ones. We have half-inch foam flex cables going to

the screen room. We pulled bundles of thirty cables around four-foot

408
radius bends in 6-inch conduit. That would seem to give a high

probability of damage. In all we pulled 270 cables, but only damaged

a couple. I have a question for Bob Lawton. You showed a TV camera

to monitor an oscilloscope and I was wondering if there is any

limitations on writing rate for that system.

Bob l^wtoji (NBS). You have me at a loss as to what the exact writing

rate is. We found one of the prime characteristics was to have it

sensitive enough that it could see the writing rate of the oscilloscope

and we chose an RCA Starlite TV camera for that application. It was adequate for the real time scopes that we had in our laboratory.

Terry McComb (NRC, Canada). A question for Bob Lawton as regards

your system accuracy of 1 percent in amplitude measurements. However,

you used a 14 bit converter. That seemed inconsistent to me. I wonder if you could explain why it was a 14 bit.

Lawton . Is the question that the 14 bit is not enough to give one part in one hundred or that it is too much? The uncertainty in the

system is not determined by the A-to-D converter. There are many

other factors which one must control, not the least of which is the

fact that it is OK to make a dc measurement but not clear how one relates that to waveforms in the high frequency range. There is a related development we have made which we call a flat pulse generator

to help us make an even more accurate determination of the voltage

level of the oscilloscope at the time frames of interest and be able

to relate those back to dc levels. So the A to D converter is not the

limiting thing there, but we find that compared to the laboratories

that have not used this high an accuracy converter, we have better

control of noise problems.

Richard Malewski (IREQ). I think that the idea of using flat straps

rather than round conductors is a generally known observation among

the measuring people. Everybody knows that it is more efficient to

use a flat strap rather than a round conductor. For the first time

we have been trying to get some quantitative information because

there is a real difference between knowing that it is better to use flat straps than round conductors and knowing what is the difference

in terms of percentages. And it is a detail, but one of importance.

I just wanted to say that we learned about how much better it is.

Hebner. Well, I want to thank you for your attendance, for your

interest, and for your comments. I want to thank all of the people who

helped make the conference a success. I would like to specifically

acknowledge and thank those people who volunteered to be session

chairmen and kept the meeting running almost on schedule for three

days. I think that we have accomplished something during the course

of these three days. I have heard discussions of transformers at

breakfast and fiber-optic communication systems at cocktails, and I have heard as much intense measurement discussion in the past two and

a half days as I could probably tolerate. I think that to most people

409
the conference was a success and the final pay-off will be when we

all go back to the institutions which sent us here and perform and

insist upon if not better measurements, at least better characterized

measurement. Thank you all for coming and I now close the session.

ft U.S. GOVERNMFNT PRINTING OFFICE : 1982 O—360-997 (21 13)

410
NBS-114A (rev. 2-8Q

U.S. DEPT. OF COMM. BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NBS SP-fi?ft

2. Performing Organ. Report No. 3. Publication Date

June 1982

4. TITLE AND SUBTITLE

Measurement of Electrical Quantities in Pulse Power Systems

Proceedings of the Workshop on Measurement of Electrical Quantities in Pulse Power

Systems held at the National Bureau of Standards, Boulder, Colorado, on March 2-4, 1981

5. AUTHOR(S)

R. H. McKnight and R. E, Hebner, Jr. , Editors

6. PERFORMING ORGANIZATION (If joint or other than NBS. see instructions)

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

FINAL

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street, City, State, ~ p Qrce Office Of Scientific Research; the Defense Advanced Research Project Agency, Directed Energy Office; the Department of Energy, Office of Inertial Fusion; the Naval Surface Weapons Center; the U. S. Army, Harry Diamond Laboratories; and the National Bureau of Standards Center for Electronics and Electrical Engineering.

10. SUPPLEMENTARY NOTES

Library of Congress Catalog Card Number: 82-600535

] Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less factual summary of most significant information,

bibliography or literature survey, mention it here)

If document includes a significant

The Workshop on Measurement of Electrical Quantities in Pulse Power Systems

addressed measurements in pulse systems having the characteristics of pulse duration

less than a millisecond, system voltages greater than 10 kilovolts, and system

currents greater than 10 kiloamperes. The presented papers were divided into

four categories: voltage measurements; current measurements; power and energy

measurements, and data acquisition. Included are discussions of applications of

conventional measurement techniques and state-of-the-art systems.

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolon s) current measurement; electrical measurements; electromagnetic pulse; fusion;

nuclear effects simulation; particle beam technology; pulse power; transients;

voltage measurements . 13. AVAILABILITY

|~Xl Unlimited

| | For Official Distribution. Do Not Release to NTIS

[~~y| Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402.

3) Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

420

15. Price

$9.50

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