NBSIR 78-881

APPLICATIONS OF THE HOMOMORPHIC TRANSFORMATION

TO TIME DOMAIN MEASUREMENT PROBLEMS

Sedki M. Riad

Norris S. Nahman

Electromagnetic Technology Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

Boulder, Colorado 80303

June 1978

IMBSiR 78-881

APPLICATIONS OF THE HOMOMORPHIC TRANSFORMATION

Sedki M. Riad

Norris S. Nahman

Electromagnetic Technology Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

Boulder, Colorado 80303

June 1978

U.S. DEPARTMENT OF COMI^ERCE, Juanita M. Kreps, Secretary

Sidney Harman, Under Secretary

Jordan J. Baruch, Assistant Secretary for Science and Technology

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director
FOREWORD

The work reported herein was accomplished by Mr. Sedkl M. Riad under a formal agreement between the Dean of the Graduate School of the University of Toledo, Toledo, Ohio, the

Dean of the Graduate School of the University of Colorado, Boulder, Colorado, and

Dr. Norrls S. Nahman of the National Bureau of Standards, Boulder, Colorado, and also

Professor, Electrical Engineering Department, University of Colorado, formerly Professor

and Chairman, Electrical Engineering Department, University of Toledo. In the course of

the work Dr. Nahman provided limited supervision and acted solely as a dissertatloii advisor.

The work was within the National Bureau of Standards Electromagnetics Measurements Graduate

Program and was partially funded by contract CST-8392, Devices and Techniques for Time

Domain Measurements, Electrical Engineering Department, University of Colorado, Boulder,

Colorado.

11
CONTENTS

Page

ABSTRACT 1

1. INTRODUCTION 1

2. THE HOMOMORPHIC TRANSFORMATION 3 2.1 Introduction 3 2.2 The Homomorphic Transformation 4 2.3 The Separation Idea 4 2.4 The Homomorphic Deconvolution 5 2.5 Computational Realization of the Homomorphic Deconvolution Transform 8 2.6 Computational Hint; the Linear Phase Elimination 9

2 . 7 Summary 10

3. THE THEORY AND APPLICATION OF THE HOMOMORPHIC TRANSFORMATION TO DECONVOLUTION

IN TIME DOMAIN LINEAR NETWORK ANALYSIS 20

3.1 Introduction 20

3.2 Time Domain Network Analysis 20

3.3 Pure Resistive Networks 21

3.3.1 Case of pure resistive line loading 22 3.3.2 Commensurate transmission lines 23

3.4 Networks with Reactive Components 24

3.4.1 Time domain transmission (TDT) 25

3.4.2 Time domain reflections (TDR) with reactive loading/unmatched

generator impedance 26

3.4.2.1 TDR measurements 26

3.4.2.2 Conventional method 27 3.4.2.3 The application of the homomorphic deconvolution method 27 3.4.2.4 Application 28

3.5 Summary and Conclusions 29

4. TIME DOMAIN DIELECTRIC SPECTROSCOPY 79 4.1 Introduction 79

4.2 Theory of Time Domain Dielectric Spectroscopy 79 4.3 The Homomorphic Deconvolution as Applied to TDDS Problems 81

4.4 Model for Debye Dielectric 81

4.5 Application of Homomorphic Deconvolution in TDDS 82

4.6 Summary and Conclusions 83

5. THE APPLICATION OF THE HOMOMORPHIC DECONVOLUTION TRANSFORM TO THE REMOVAL OF SCATTERING COMPONENTS FROM ANTENNA TEST RANGE DATA 90

5.1 Introduction 90

5.2 The Homomorphic Deconvolution as Applied to Antenna System 91

5.3 The Antenna System 91

5.4 The Application 92

5.5 Summary and Conclusions 93

iii
CONTENTS (Continued)

Page

6. TRANSFER NETWORK IDENTIFICATION (MODELING) OF FEED-THROUGH SAMPLING-HEAD 118

6.1 Introduction 118

6.2 Theory and Construction of Sampler 118

6.3 Modeling of Physical Structure 119

6.4 The Biconical Cavity 119

6.5 Sampling Diodes r 122

6.5.1 Static measurements (I and R ) 122

s s 6.5.2 Dynamic measurements (C^, L^, and C^) 123

6.6 The Sampling Pulse 125

6.7 Structure Step Response 128

6.8 Sampling Process 128

6.9 The Effect of the Diode Bias on the Sampling-Head Step Response 130

6.10 The Deconvolution Routine 130

6.10.1 The homomorphic deconvolution method 131

6.10.2 An application 131

6.11 Summary and Conclusions • 132

7. SUMMARY AND CONCLUSIONS-^ ~ 179

8 . ACKNOWLEDGMENT 181

APPENDIX A -- THE z-TRANSFORM AND THE DISCRETE FOURIER TRANSFORM 182

APPENDIX B ~ COMPUTER SUBROUTINES HDT AND HDF 184

APPENDIX C ~ COMPUTER PROGRAMS TDNA AND SMPL 187

C-1 Program TDNA 187

C-2 Program SMPL • 189

C-3 Subroutines Used in TDNA and SMPL 191

REFERENCES 199

iv
APPLICATIONS OF THE HOMOMORPHIC TRANSF0R21ATION

TO TE'IE DOMAIN MEASUREMENT PROBLEMS

Sedkl M. Rlad and Norris S. Nahman

This report presents a study of the theory and application of the ho~omorphic transformation to deconvolution problems occurring in time domain

measurements in the picosecond time domain. A homomorphic deconvolution transform was developed and applied successfully to remove the time-windowing restriction required in many time domain measurements. Examples were presented

including problems in time domain analysis of linear networks and dielectric

spectroscopy, and scattering and multiple reflection in antenna (radiation)

systems were considered and treated. Also considered was the development of a model for a 28 picosecond resolution feed-through sampling-head, and the

model's step response was computed. Simulation studies were performed using

typical input waveforms and the oscilloscope model. The homomorphic transformation was used to deconvolve the model's impulse response from the simulated

output. Comparison of the deconvolved output waveforms with the input waveforms

showed agreement within the accuracy of the sampled-data simulation.

Key words: Antenna scattering; deconvolution; dielectric spectroscopy;

homomorphic transformation; modeling of sampling gates; signal processing; time domain measurements; time domain ref lectometry

1 . INTRODUCTION

This report is concerned with the study of the theory and application of the

homomorphic transformation to deconvolution problems occurring in time domain spectroscopy

and scattering. Typical time domain problems that fall under this category include the

following: dielectric and magnetic spectroscopy, time domain ref lectometry and transmission (network spectroscopy), and wave scattering and multiple reflections in antenna

problems. Conventionally, time domain measurements methods are based upon and limited by

time-windowing to select the relevant or desired waveform epoch.

In this report a new method is developed based upon the mathematical homomorphic

transformation which removes the time-windowing restriction required in the usual time

domain measurement method. This allows the use of overlapping time vrindows which effectively removes the time-window restriction and thus engenders a new method for time

domain spectroscopy, and in general, a new method for time domain measurements in the

picosecond time domain [1]

Furthermore, because time domain representations have the property of directly

exhibiting reflected or scattered contributions, this method has significant impact in the

broad class of problems falling under the category of scattering problems which are encountered widely in electrical engineering and in physics [1]

Specifically, this report develops four problems, or applications, which are encountered

in various forms in time domain spectroscopy:
1. Deconvolution of incident and scattered components in the time domain

analysis of linear networks

- 1 , 2. Removal of the time windowing restriction caused by a finite-sized specimen

in time domain spectroscopic studies of dielectrics.

3. Removal of scattering components from antenna test range data.

4. Determination (deconvolution) of the excitation waveform from the observed

" sampled-data response of a picosecond domain sampling oscilloscope.

In summary, this report explores in depth the theory and application of the homomorphic transformation to scattering problems using the above topics as examples, including

experimental and computer data reduction studies.

This report is organized into seven chapters and three appendices. The present

introduction being the first chapter.

Chapter 2 will be devoted to the study of the theoretical aspects of the homomorphic

transformations, and in particular, the homomorphic deconvolution transformation. Also the

computational realization of the homomorphic deconvolution transformation will be given.

In chapter 3 the homomorphic deconvolution transform developed in chapter 2 will be

applied to various time domain linear network analysis problems. The applications will

cover both time domain ref lectometry and transmission measurements for pure resistive

networks (e.g., commensurate transmission lines network), as well as networks with general

(reactive) reflective elements (discontinuities)

Chapter 4 will present the theory and limitations concerning time domain dielectric

spectroscopy. The homomorphic deconvolution transformation will be applied to remove the

restrictions of the conventional method.

In chapter 5 the homomorphic deconvolution transform will be applied to another class

of time domain scattering problems: the removal of scattering components from antenna

test range data.

Chapter 6 will be concerned with the transfer network identification (modeling) of

feed-through sampling head. The theory and construction of the sampling-head will be

reviewed. Various experimental and computational steps involved in the modeling process

will be presented. The model will be used to compute the sampling-head step response. A

deconvolution routine is to be developed in order to deconvolve the sampling-head's

response contribution from the observed waveform on the oscilloscope to get an estimate of

the actual waveform at the input port of the oscilloscope.

Chapter 7 will contain the summary and conclusions of the work presented in this report.

Relevant detailed mathematical developments and computer programs are referred to in

the text and presented in appendices A through C.

-2-
2. THE HOMOMORPHIC TRANSFORMATION

2.1 Introduction

It is often encountered in time domain measurements, as well as in other various

applications, that an observed signal or a waveform x(t) can be represented by the convolution of two (or more) components x^(t) and X2(t); that is,

GO x(t) =

/ x^(t') X2(t-t') dt' (2.1a)

—CD

or using the simplified notation

x(t) = x^(t) * X2(t) (2.1b)

where the (*) denotes the convolution process. In the frequency domain, eq. (2.1) takes

the form

X(ej'') = X^(ej") • X2(e>) (2.2)

where X(e-^"), X^(e-''^), and X2(e-''^) are the Fourier transforms of x(t), x^(t), and X2(t),

respectively, all complex functions of e-''^. Some of the examples of time domain convolutions are (a) the reflected signal on a transmission line which is the convolution of the incident signal and the impulse response

of the reflecting discontinuity and (b) the observed signal using an oscilloscope which is

the convolution of the measured signal (at the input gate) and the oscilloscope's impulse

response

In order to separate the components of a convolution, a deconvolution (separation)

process is needed. In the past several deconvolution methods were developed for various

convolution problems; they all required the knowledge of either one of the convolution

components, and in some cases, information about the signals being analyzed. A fairly new

method of signal processing, the homomorphic transformation [2,3,4,5], will be described

in the present chapter and will be used throughout the following chapters in deconvolving

time domain signals. The new method does not require the knowledge of any of the convolution components and suits the class of problems where the frequency domain forms of the

convolution components have substantially different rates of variation. As will be shown

later, the time domain ref lectometry and spectroscopy problems do yield to the new

deconvolution method.

Since all signal-processings that will be encountered in the deconvolution process

will be carried out using digital computers, the analysis will be presented in the discrete

(sampled-data) form.

-3-
2.2 The Homomorphic Transformation

A homomorphic system H, figure 2.1, is a nonlinear system, which obeys the generalized

principle of superposition [2,3,4]; i.e.,

H[x^(n) 0^ x^Cn)] = H[x^(n)] 0^ H[x2(n)] (2.3)

and

H[c : x^(n)] = c i H[x^(n)] (2.4)

where x^(n) and X2(n) are two input sequences to the system H, n is the sequence variable,

c is any scalar quantity, 0^ (or 0^) denotes a rule for combining inputs (or outputs) with

each other (e.g., addition, multiplication, convolution, etc.), and : (or \) denotes a rule for combining input (or output) with a scalar. The class of linear systems is a special case for which 0^ and 0^ are addition and : and i are multiplication. Such

systems are said to have an input operation 0^ and an output one 0^.

The canonic representation of homomorphic systems is shown in figure 2.2. Both

2r

(2.4),

systems , and D^^ obey the generalized principle of superposition, eqs . (2.3) and

i o

The input operation for is 0^ while the output one is + (addition) ; the system L

i -1

is a linear system, and the system D_ transforms from + to 0 .

•^ 0 o o

' - ^ - oi ; : IV - ; : ; 2.3 The Separation Idea

Referring to figure 2.2 and letting x(n) denote the combination of the input sequence

Xj^(n) and X2(n), it can be written that

[x(n)] = Dq [x^(n) 0. X2(n)] = [x^(n)] + [x2(n)]

i i i i

' ' = x^(n) + X2(n)

= x(n) (2.5)

L[x(n)] = L[xj^(n) + X2(n)] = L[x^(n)] + L[x2(n)]

= y(n) (2.6)

DQ^[y(n)] = DQ^[yi(n) + y2(n)] = DQ^[y3^(n)] 0^ DQ^[y2(n)]

o o o o

= y^Cn) 0^ y2(n)

= y(n) . (2.7)
Now, if it is required to recover one of the input signals (e.g., x^(n)) from the combined

sequence x(n), the linear system L must have the filtering property

L[x^(n) + x^Cn)] = x-j^(n); (2.8a)

i.e.

,

y(n) = x^(n). (2.8b)

Then, choosing

Dq = % (2.9)

o 1

eqs. (2.7) and (2.5) yield

y(n) = DQ-'-[Xj^(n)] = x^(n) . (2.10)

i

In conclusion, the separation of the combined sequences x^(n) and X2(n) depends on the success in separating their transforms x^(n) and X2(n) , which depends in turn on the

properties of the signals involved.

2.4 The Homomorphic Deconvolution

As mentioned earlier in section 2.1, the deconvolution is the process of separating

convoluted signals from each other. The homomorphic deconvolution is then the use of a suitable homomorphic system (of transformations) to separate (deconvolve) a convolution.

The input and output operations of a homomorphic deconvolution system are the convolution

operation (*) , according to eq. (2.9). The canonic form of a homomorphic deconvolution

system is shown in figure 2.3. In the figure, the transform converts the convolution

operation into an addition. The transform D^''" is the inverse transform of D^^. In con- structing the system to do its function, there are two useful mathematical operations

that will be used

a. The z-transform z[ ], appendix A, converts a convolution into a product,

eq. (A. 13). In other words, the z-transform is a homomorphic transform with (*)

and (•) as the input and output operations, respectively.

b. The complex logarithm operator log converts a product into a sum; i.e., the log

is a homomorphic transform with (•) and (+) as the input and output operations,

respectively.

The simplest form for the transform can then be the cascade combination of the z- transform and the complex logarithm, figure 2.4a, but if it is preferable in the separation

process to deal with sequences rather than their z-transf orms , the addition of an inverse

z-transform z ^[ ], eq. (A. 3), which is a linear one, will not alter the output operation,

figure 2.4b.

-5-
Studying the signal flow in the constructed system D^^ of figure 2.4, with x(n) as the

convolution of two sequences Xj^(n) and X2(n), the following relations are obtained:

X(z) = z[x(n)] (2.11a)

= z[x^(n) * x^Cn)] (2.11b)

= z[x^(n)] • z[x2(n)] (2.11c)

= X^(z) . X^(z) (2. lid)

X(z) = log[X(z)] (2.12a)

= log[X^(z) • X2(x)] (2.12b)

= log[X^(z)] + log[X2(z)] , (2.12c)

X^(z) + X2(z) (2.12d)

x(n) = z'^[X(z)] (2.13a)

= z"^[X^(z) + X^(z)] (2.13b)

= z"^X^(z)] + z"^X2(z)] (2.13c)

= x^(n) + X2(n) . (2.13d)

The reason for denoting the sequence variable in eq. (2.13) by n is to distinguish the new domain from the original n-domain. This n-domain is a nonphysical domain; it is a pure

mathematical one or what is to be called the hypothetical n-domain. If the n-domain is the

time domain, then the n-domain is to be referred to as the hypothetical time domain. The

signal x(n) in the hypothetical n-domain is historically named the cepstrum (reversing the

order of the first four letters of the word spectrum).

The separation process will then be achieved using linear filters in the n-domain to split x(n) into its components x^(n) and X2(n). The filtration problem will be greatly

simplified if x(n) is real. Assuming that both x(n) and x(n) are real stable sequences is

not in fact a restriction since these are the properties of the majority of signals encountered

in this work.* Besides, simple mathematical operations can change the stability conditions

of a given signal. For example, exponential weighting can be used to secure stability as being discussed in appendix A. This assumption implies that the regions of convergence of

both X(z) and X(z) must include the unit circle, see appendix A. On the unit circle z = e~''^, X(z) can be expressed as X(e-''^)

,

X(ej") = X^(e^-) + jX^(ej-)

^^.14)

*Here a stable signal is defined as one whose z-transform contains no poles exterior to or on the unit circle. Note that this definition corresponds to a stable signal as being

defined as one whose Laplace or s-transform contains no poles on the imaginary axis or

in the right-hand plane.

-6-
where the subscripts R and I refer to real and imaginary components of a complex quantity.

Since x(n) is assumed to be real, X_,(e-''^) and X (e-^*^) must respectively be even and odd

periodic functions of w with period 2tt. Both functions must be analytic (continuous

functions of oo) to meet the stability requirements. Since X = log X by virtue of eq. (2.12),

the analyticity of the complex logarithmic function is to be studied.

The complex logarithmic function log is defined as

log X = log|x] + j arg[X] (2.15a)

- ir < arg[X] <_ + n. (2.15b)

Using eqs. (2.12) and (2.14), together with the above definition, the real and imaginary

components of X(e'^'^) are

X^(ej'^) = log|x(e^'^)| (2.16a)

and

X^(ej") = arg[X(e^'')] . (2.16b)

The real part, eq. (2.16a), is analytic as long as the z-transform X(z) does not have any

zeros on the unit circle. However, the exponential weighting method, appendix A, can be

used to move any zeros lying on the unit circle to make eq. (2.16a) analytic. The imaginary

part, eq . (2.16b), is not a continuous function according to eq. (2.15b); this does have

two undesired effects:

a) The eq. (2.12c) will not (always) be true. Using eq . (2.14), eq . (2.12) becomes

log|x| = log|X^| + loglx^l (2.17a)

arg[X] = arg[X^] + arg[X2] (2.17b)

where the three arguments in eq. (2.17b) are all defined in the region -tt to +tt. The relation of eq. (2.17a) is always true, but eq . (2.17b) is not because of the

restrictions on the arguments.

b) X(z) will not (always) be a valid z-transform because of the noncontinuity of the

imaginary component of X(= log X), thus invalidating eq. (2.13).

The solution to the complex logarithm problem is simple; it is to replace the discontinuous complex logarithm by the continuous complex logarithm clog defined as

clog[X] = log|x| + j carg[X] (2.18a)

carg[X] = arg[X] + 2mTr (2.18b)

where carg is the continuous argument function. This new function is formed by recon- structing the argument function to be a continuous one by addding (or subtracting) integer

multiples of 2-n , figure 2.5.

-7-
In conclusion, for the system D^^ to function properly (i.e., convert the real stable

convolution x(n) into a real stable sum x(n)), the constructed clog function must be used

instead of the log function; and the real and imaginary components of both X(e-''^) and

X(e'^'*^) must be respectively even and odd continuous periodic functions of cd with period Zir. The inverse system D^"'' should perform the reverse processes of the system D^; i.e.,

convert an addition at the input to a convolution at the output. D^''" is constructed by

reversing the order of the transforms of replacing each transform by its inverse,

figure 2.6. The system D^''' does not have any ununiqueness problems. Since the system

D^''' is the inverse transform of D^,

D;^D^[x(n)]] = x(n). . ^

.

Since both x(n) and x(n) are assumed to be real and stable, it follows that both y(n)

and y(n) are both real and stable. Thus the regions of convergence of both Y(z) and Y(z)

must include the unit circle.

2.5 Computational Realization of the Homomorphic Deconvolution Transform

In most physical situations the observed signals have to be dealt with numerically

rather than analytically. The digital computer is often used to perform various kinds of

numerical processing on such signals. For the separation of physical convolution signals,

a digital computer version of the homomorphic deconvoluting transform has to be derived.

In section 2.4, it was decided that the z-transforms X(z), X(z) , Y(z), and Y(z) have

to include the unit circle in their regions of convergence. Thus the inverse z-transforms

can then be calculated using the unit circle as the integration contour, eq. (A. 3). Accordingly, the evaluation of the above mentioned z-transforms is only needed on the unit

circle; i.e., it is employed to calculate the signal Fourier transforms rather than the

z-transforms, eq . (A. 6). Because of the discreteness and finiteness required in numerical

computational methods, the input signal has to be used in its discrete version. The

discrete Fourier transform, figure 2.7, shows the computational realization of the homomorphic

deconvoluting transform, in which the signals are related as follows:

X(k) = XCeJ'")

N-1

w = 2Trk/N

-j (2TTk/N)n

=

I x(n)e-J^"^'^'^^" (2.19)

n=0

X(k) = clog[X(k)]

= log|x(k) I + j carg[X(k)], (2.20a)

carg[X(k)] = arg[X(k)] + m2TT. (2.20b)

-8-
Here m may be zero or a positive or negative integer, such that carg[X(k.)] is continuous.

kh =1 Y X(k)ej(2"^/^>" (2.21)

k=0

y(n) = L[x(n)] (2.22)

Y(k) = Y^(n) e-J^'^^/")" (2.23)

n=0

Y(k) = exp[Y(k)] (2.24)

y(n) =iT^(k)e^^'"^^''^" • (2.25)

k=0

It has to be noted, that the input sequence x(n) is the convolution of two sequences x^(n)

and X2(n) both of finite number of samples N; consequently, the cepstrum x(n) is the sum of

the two contributions x^(n) and X2(n) due to x^(n) and X2(n), respectively. The output of

the linear filter y(n) should be equal to either of the components Xj^(n) or X2(n) to yield

a system output y(n) which correspondingly equals either Xj^(n) or X2(n). Also, it is to be

noted that all the sequences involved in the transform equations, eqs. (2.19) through

(2.25), either in the n, k, or n-domains are each of a finite number of samples N, and they

each satisfy the relation

h(m) = h(m + rN) ; r integer; (2.26)

i.e., all the computations need to be done only for sequence variables ranging between 0 and N-1.

The system of figure 2.7 was simulated by the computer subroutine HDT given in

appendix B. In cases where the input and output signals to the deconvolution routine are represented by their discrete Fourier transforms, the subroutine HDF is to be used. HDF is

the same as HDT except that it does not contain the first DFT and the last DIET.

2.6 Computational Hint; the Linear Phase Elimination

Let the sequence x(n), the input to the homomorphic deconvolution transform of

figure 2.7, be the result of delaying another sequence s(n) by n^ samples; i.e.,

x(n) = s(n-n^). (2.27)

-9-
Following the transform equations, eqs . (2.19), (2.20), and (2.21), it can be written

that

X(k) = S(k)

-j(2Trk/N)n^

e (2.28)

X(k) = S(k) - j(27rk/N)n^ , (2.29)

and

x(n) = s(n) + (-1)

n+1

n /n . o (2.30)

The second term in eq . (2.30) is of the form of a decaying oscillation. This term is

called the linear phase term because it is due to the linear phase component in the Fourier

transforms, eqs., (2.28) and (2.29). Assuming that it is desired to perform linear filtering

on s(n), the linear phase term must be removed. Said another way, s(n) contains the desired

information while the addition of the linear phase term obscures this information. Accordingly, the linear phase term of x(n) must be removed. Figure 2.8 illustrates how the

linear phase term distorts the information s(n) contained in x(n)

The existence of such a linear phase term also contradicts the requirements of having

odd periodic continuous phase (argument) function with co. This linear phase component has

to be eliminated, and the best place to perform the elimination is right after taking the

clog of the DFT of the signal. Figure 2.9 shows the linear phase elimination process.

This chapter started with a review of the convolution relations in both the time and

frequency domains, and the need for deconvolution (separation) techniques.

The general class of the homomorphic systems was defined, and their properties were

stated showing how they can be used for signal separations. In particular, the homomorphic

deconvolution transform was developed. The related problems, such as the nonanalyticity of

the complex logarithmic function and the effect of linear phase component, were presented

and discussed, and solutions to such problems were given.

The computational realization of the deconvolution transformation was made possible in

the form of digital computer subroutines HDT and HDF.

In the next chapters, the new deconvolution method will be put into use in deconvolving

various kinds of time domain signals, such as those observed in time domain spectroscopy

measurements

V

2 . 7 Summary

-10-
x(n) = x^(n)0.X2(jn

1 (n)

H y(n) = H[x(n)]

'= H[x^(n)]0^H[x2(n)]

Figure 2.1 A Homomorphic System H with 0^, and 0^

as the Input and Output Operations,

Respectively.

x(n)

+ + + + 0,

0

0

! L y(n)

y(n)

Figure 2.2. Canonic Representation of Homomorphic

Systems

x(n)

+ +

x(n)

+ +

y(n) y(n;

Figure 2.3. The Canonic Form of a Homomorphic

Deconvolution System.

-11-
* » • +

z[ ] log

x(n) X(z) X(Z)

a. The Output Signal is a z-Transform.

*

z[ ] log

+ +

x(n) X(z) X(z)

b. The Output Signal is a Sequence.

Figure 2.4. Possible Forms for Construction of the

Transform , a Homomorphic System with

(*) as the Input Operation and (+) as

- the Output Operation.

-12-
arg[X(ej*")]

TT M ^ V

7T W

71 U)

carg[X(e^'")]

Figure 2.5. The Reconstruction of the Argument

Function arg to Create the Continuous

carg Function.

-13-
y(n)

z[ ] EXP[ ]

* * z-\ ]

1 Y(Z) y(n)

Figure 2.6. The Transform D

-1

-14-
LINEAR

FILTER

L[ ]

y(n) DIFT[ ] EXP[ ] DFT[ ]

Figure 2.7. The Computation Realization of the

Homomorphic Deconvolution Transform.

DFT = Discrete Fourier Transform.

DIFT = Discrete Inverse Fourier

Transform

CLOG ^ Continuous Complex Logarithm

Function.

-15-
-16-
-17-
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M-l

H-t rH w D

>+-l 0)

tn El D

00u

=) •H

fa
N/2

Figure 2.9. The Elimination of the Linear Phase

Component

-19-
3. THE THEORY AND APPLICATION OF THE HOMOMORPHIC TRANSFORMATION

TO DECONVOLUTION IN TIME DOMAIN LINEAR NETWORK ANALYSIS

3.1 Introduction

In chapter 2, the homomorphic deconvolution transform was developed. As it was shown,

the separation (deconvolution) idea was based on the property of this transform to convert

a convolution of two sequences (waveforms) into a sum of their contributions in a hypothetical time domain called the cepstrum. Separating filters are then to be applied to the

cepstrum sequence to select only one contribution at a time. The separation can be realized

easily if the two cepstrum contributions were discernible from each other; i.e., each one can be viewed in a separate window, in which case simple (ideal) bandpass type (mathematical)

filters can be used for the separation process.

In the present chapter, a class of problems that meet the above mentioned requirement

for easy separation will be treated; this class of problems is the analysis of linear

microwave networks using time domain techniques.

3.2 Time Domain Network Analysis

The time domain method of network analysis is based upon observing the reflections (or

transmission) of an incident time domain waveform back from (or through) the network to be

analyzed. The observed signal contains information on both the incident signal and the

networks impulse response. The typical experimental setups for both the reflection and

transmission cases are given in figure 3.1. The method of observing the reflected waveform

is commonly called the TDR (time domain ref lectometry) method; on the other hand, the

abbreviation TDT will be assigned to the time domain transmission measurement method. In TDR, figure 3.1a, the observed reflected waveform can be viewed as the convolution of the

incident waveform and the reflection impulse response of the network. While in the TDT,

figure 3.1b, the transmission waveform is the convolution of the incident waveform and the

transmission impulse response of the network. The general nature of the impulse responses

of the microwave networks that will be considered in this chapter is of the form of a pulse

train, because of discretely distributed discontinuities along the network transmission

line. The frequency domain form of such impulse responses do have relatively rapid variations (with frequency) compared to the rates of variation occurring in the frequency domain

forms of most test (incident) waveforms commonly used in TDR or TDT measurements. Consequently, the application of the homomorphic deconvolution transform will result in clustered

groups of components in which the two distinct contributions will be due to the network

impulse response and the incident waveform, respectively; and accordingly they will be

easily separable. Figure 3.2 shows the homomorphic deconvolution transformation as being

applied to a typical time domain waveform that could be encountered in either a TDR or a TDT measurement

-20-
The microwave networks that will be considered in the next two sections of this

chapter will be classified into the following two classes

(1) Pure resistive networks, i.e., with pure resistive reflections (discontinuities),

and this will be considered in section 3.3.

(2) Networks with reactive discontinuities, which will be dealt with in section 3.4.

The reason for this classification is, as will be seen later, that a pure resistive network

has a train of delta functions as its impulse response in comparison with reactive reflections, where the impulse response contains a train of pulses whose duration and time

constants increase with an increase in the reactive parts of the discontinuities. This

will appear in the cepstrum domain in the form of more separability for the resistive

network as compared to reactive ones.

3.3 Pure Resistive Networks

In this section two types of resistive discontinuity problems will be considered:

(1) pure resistive loading (discontinuities) along a lossless transmission line, figure 3.3

and (2) the case of commensurate transmission lines, figure 3.4, which is a cascade of

lossless transmission lines, each one having different electrical parameters (Z^, y) . In practice, these two considered cases may include reactive components in their physical

realization; but such reactive components are always neglected in the idealized case, and

the networks are treated as purely resistive ones. However, the extension to cases with more general reflections including reactive parts will be treated in the next section. , In this section the time domain ref lectrometry (TDR) method will be used. The general

form of the reflective impulse response h'(t) of a microwave resistive network is given by

h'(t) =

I A^6(t-T^) (3.1)

n=0

where t is the time variable and both A and x are constants determined by the network

n n

^ parameters. Assuming an incident waveform x(t) at the input terminals of the network, the

reflected waveform y'(t) would be the convolution of x(t) and h'(t); i.e.,

y'(t) = x(t) * h'(t). (3.2)

The observed waveform y(t) at the input gate is the sum of both incident and reflected

ones ; i.e.,

y(t) = x(t) + y'(t)

= x(t) + x(t) * h'(t)

= x(t) * (5(t) + h'(t))

= x(t) * h(t) (3.3)

-21-
where

h(t) = 6(t) + h' (t).

In the frequency domain the signal h(t) transforms into H(e-''^) where

HCeJ'^) = 1 + I A e n=0

00 '00 = 1 + 5] A COS 0) T - j A sin w x (3.5)

n=0 " n=0 " "

which is clearly seen to be oscillatory in nature (with frequency). On the other hand

X(e-''^) for most physical signals is non-oscillatory; i.e., slowly varying with frequency

compared with H(e~''^). The application of the homomorphic deconvolution can then be

expected to separate the two components x(t) and h(t) from their convolution.

In the following the two above described cases of resistive networks will be considered.

In each case two examples will be considered and developed; the first of which will be

simple (low number of discontinuities) such that the mathematical verification of the

results can be followed without too much confusion. The second example containing a large

number of discontinuities will be presented in terms of the results of applying the homomorphic

deconvolution in order to demonstrate the ability of the method to handle such complicated

cases.

In the numerical computations for the examples presented below, 1024 samples in both

time and frequency domains were used. The time and frequency sampling intervals were

2.45 ps and 0.4 GHz, respectively.

3.3.1 Case of pure resistive line loading

The general form of network to be considered in this part is given in figure 3.3.

However, as mentioned earlier, only a simple case will be considered as far as the mathematical details are concerned; this is the case of two discontinuities shown in figure 3.5. Two resistances and are connected across a lossless transmission line having a characteristic impedance R^ at the distances and shown in the figure. Denoting

the reflection coefficients at the discontinuities R^^ and R2 as and respectively,

the reflective impulse response of the network is derived in figure 3.5b to be

h'(t) = p^6(t-2T^) + p^(l+p^)^ 6(t - (21^+2X2)) + .... (3.6)

Upon substituting for p^, P2, x^, and X2 by their respective values -0.2, -0.75, 127 ps, and 84.67 ps that were used in the simulated application, this impulse response becomes

h'(t) = -0.2 6(t-3x) - 0.48 6(t-5x) -0.072 6(t-7x) - (3.7)

-22-
where t = 84.67 ps . Figure 3.5c shows the impulse response h(t) = 6(t) + h' (t) for this

specific case. Figure 3.6 shows the time domain waveform Sg(t) of the generator used, while figure 3.7

shows the corresponding frequency domain form. The incident waveform x(t) is given by

e (t)

x(t) = -^2~ (3.8)

since Rq the generator internal resistance is matched to the line resistance. Thus x(t) can also be represented by the figures 3.6 and 3.7 scaled by a factor of one half.

Figures 3.8 through 3.14 show various processing waveforms throughout the steps of

the homomorphic deconvolution transform. Figures 3.8 and 3.9 show the observed waveform

at the input gate of the network in both time and frequency domains, respectively. Figure

3.10 shows the resultant cepstrum and the linear filters to be used for the separation

process. Figures 3.11, 3.12, 3.13, and 3.14 give the time and frequency forms of the

deconvoluted components x(t) and h(t), respectively. The obtained forms of x(t) and h(t)

do coincide with the original form to a great extent. The erroneous oscillations observed

in the deconvoluted impulse response is in fact due to sampling errors; i.e., finite

number of samples and finite observation time (window)

The figures 3.15 through 3.19 show another example, but a more complicated one with

six discontinuities along the line. Figure 3.15 shows the network considered and various

values of various network parameters. The generator signal used is the same one as given

in figures 3.6 and 3.7. Figure 3.16 shows the convolution y(t), while figure 3.17 shows

the corresponding cepstrum. The deconvoluted x(t) and h(t) are given in figures 3.18 and

3.19, respectively.

3.3.2 Commensurate transmission lines

The general form of a commensurate transmission line network was given in figure 3.4.

Again, two examples will be considered in this part, one example with two discontinuities

and the other one with six discontinuities. The same generator signal that was used in

section 3.3.1 will be used in this section.

The two discontinuity example is shown in figure 3.20. Figure 3.20a gives the

network; it is a cascade of two lines having the following parameters:

-23-
Characteristic

Impedance

Line 1 Line 2 \ + jO + jO

2 Propagation n • / r. . / Constant -^1

Length . ^1 ^2

s One Way „ , „ ,

• Travel Time ^1 = V^l ^2 = V^2

The exiting generator has a matching impedance R^. Figure 3.20b shows the multiple

reflections pattern. The reflective impulse response as can be written from this figure is

h'(t) = p6(t-2T^) - p(l-p2) 6(t - (2x^+2x2)) - p3(l-p2) 6 (t - (2x^+4x2)) - .... (3.9)

While h(t) = 6(t) + h' (t) is shown in figure 3.20c for the specific case that is to be

simulated on the computer, in which

p = 0.333, . ,

x^ = 127 ps,

and

X2 = 101.6 ps, .. '

'

this h(t) is given by

h(t) = 6(t) + 0.333 6(t-3x) - 0.2963 6(t-5.4x) - 0.0329 5(t-7.8x) - (3.10)

Figures 3.21 through 3.27 show the processing waveforms throughout the steps of the

homomorphic deconvolution transform. Again, figures 3.25 and 3.27 give the deconvoluted

x(t) and h(t) which are demonstrated to be in complete agreement with the original forms

for x(t) and h(t). On the other hand figures 3.28 through 3.32 show the six commensurate

lines case.

3.4 Networks With Reactive Components

In the present section two additional applications will be introduced beyond the

generalization to more general discontinuities along the line. Section 3.4.1 will be

devoted to the consideration of a TDT type measurement, while section 3.4.2 will deal with

the effects of an unmatched source impedance in a conventional TDR measurement.

-24-
3.4.1 Time domain transmission (TDT)

Figure 3.1b, earlier in the chapter, showed the experimental setup for a typical TDT

measurement. The example that will be considered in this section is the following:

two impedance discontinuities and were connected across a line having characteristic

impedance and a propagation constant at distances d^ and as shown in figure 3.33.

A generator of matched impedance excites the network; ^g(t) is the generator time domain

waveform. The aim of this specific problem was to see how the TDT waveforms would yield to the homomorphic deconvolution transform.

In the numerical computations for the present example, 1024 samples in both time and

frequency domains were used. The time and frequency sampling intervals were 4.9 ps and

0.2 GHz, respectively.

The values of the network parameters that were used in this specific example were as follows

line: 7 mm precision coaxial line (50 H) [6] Z^: Capacitive, = 0.3 pf

Z^: Capacitive, = 0.1 pf

d2 : 3.54 cm

The waveform of the generator ^gCt) is given in figure 3.34. The network was simulated on

a digital computer, and the resultant TDT waveform at the outout terminals e (t) is given

o in figure 3.35. The frequency domain form E^Ce-'"') of e^(t) is given by the relation

E (e^^) exp[-Y^(o:)- (d +d )] [1+p (O] [l+c,(oj)]

° 2 [l-p^(ai)P2(!^)expr-2Y^(uj)d2)]

where E (e-^'^) is the frequency domain form of e (t), and c^C^) and c-C-) are the reflection

g g 1 coefficients corresponding to the impedance discontinuities Z^(^) and Z^{^) , respectively.

The magnitude of E^Ce-'") is given in figure 3.35. The homomorphic deconvolution transform,

figure 3.2, is to be applied on the output signal to separate its convolution components.

As was mentioned before, the filtration in the cepstrum domain separates two components

that have substantially different rates of variation in their frequency domain forms. By

examining eq, (3.11) it can be concluded that the slowly varying component E (e"'^) would

OS

be

E (ej")

E (e^") = exp[-7 (^)-(d^+d„)] [1+P,(-.)] [l+=,(^)] (3.12)

OS ^ o L I 1 I

which is the first path signal through the network, while the rapidly varying one ^^^(e-'")

is the multiple reflection effect,

E (e^'^) . (3.13) or

[l-p^((jo)p2(w) ] exp -2Y^(aj)d2

-25-
These eqs. (3.12) and (3.13) can be used to determine up to two unknown system parameters,

provided that all other network parameters are known. To show this, eq. (3.13) will be

used to calculate P2(<Jj) assuming that p^(e-''^) is known; the values of P2((jj) are to be used

to calculate the corresponding impedance Z2(w).

The above mentioned steps, in the current section, were followed, and the results are shown in figures 3.37 through 3.42. Figure 3.37 gives the cepstrum of e^(r). Figures 3.38

through 3.41 give the slowly and rapidly varying components in both frequency and time

domain forms, respectively. The resultant values for E^^(e-'^) were used for the calculation

of Z^iiii) as indicated earlier. The computed values for Z^(ii>) showed a pure capacitive

impedance, and the capacitance is plotted in figure 3.42. The figure shows erroneous oscillation about the actual given value. These error oscillations are again due to sampling errors. Part of the errors are also due to the imperfect separation in the

cepstrum domain because of the extended long tails due to reactive reflections on the line.

3.4.2 Time domain reflections (TDR) with reactive loading/unmatched generator impedance [7,8]

In this case the TDR measuring technique will be presented in its conventional form,

showing the limitations of the method. Then it will be shown how the homomorphic

deconvolution transform helps in offsetting these limitations.

3.4.2.1 TDR measurements

Figure 3.43 shows the typical TDR experimental setup for measuring the reflection

coefficient of the line termination p^(aj). The general procedure is to record the displayed signal for the termination under test and two other reference terminations, usually

a short circuit and a Z^(a)) termination.* The complex frequency domain forms of the three

recorded signals are given by

Z (o)) E^^(eJ'') = E (e^") • ^ • T((.) (3.14)

dM'

Z (co) + Z (uj) o e

E,.(ej'^) = E (ej'^) 1 - exp f-2Y (o)) ^^^^ (3_^5)

Z (to) + Z (w) 1 + p (w) •exp(-2Y(a3)£}

o g g

Z(i^) 1 + p (to) exp(-2Y(u))£}

E^Ae^"") = E (eJ'^) ^ h T(to) (3.16)

dL'

Z (03) + Z (o)) 1 - P,((o)p (to) exp f-2Y(to)£)

o e L " ' ^

corresponding to Z (to), short circuit, and Z (tjj) terminations, respectively.

O Li One cannot solve the above three equations, as they are, to obtain p^((d),

*For an exciting waveform which becomes of absolutely constant value before the reflections

-26-

arrive, the Z measurement is not necessary,

o
3.4.2.2 Conventional nethod

The conventional method for evaluating (-) is to consider only the TDR signals

occurring in the tirae window 0 < t < 41, where I is the travel time on the line of length

I. In this case ecs. (3.11), (3.12), and (3.13) will take the simplified foms

"dM,4T

Z (_)

(eJ*^) = E CeJ'") • 5 . T(cj) (3.17)

^ Z^(_) + Z(^)

o g

Z (:j) E^c ATfe-"") = S (e^'') • • T(-) • [1 - (l-H)„(«a)) exp (-2y (o) l) ] (3-18) "^'^^ S Z (O - 2 (.) S

7 C

E.. ^^ie^"") = E^(e^-) ^ • T(u) • [1 + p. (to) [l-^pC^)] exp (-2y (w) ] (3.19)

""^'"^ ^ Z (-) ^ Z (-) ^ ^

o g fron which (_) can be deteriiined;

E,. ,_(-) - E^,^, _(w)

pjcj) = - ^=^^^^^^

. (3.20)

This tiethod vorks only if the second and subsequent reflections do not overlap the

first one. Figure 3.44 shows two cases of TDR signals: the non—overlapping reflezticns

where the conventional method works, and the overlapping reflections in which the conventional

method fails

3.4.2.3 The application of the ho-oncrphic ceconvolution method

In the homomorphic deconvolution method, see figure 3.2, the ccnplete forr.s, ecs. (3.11), (3.12), and (3.13), are used to obtain not only (^) but also c^(!j)), 7(w), and the

product E^(e^'^)T(ii)) . The process is begun by first obtaining the differences of

eq. (3.15) - eq. (3.14) and eq. (3.16) - eq. (3.14) which are

^S^^'") = ^ds(^''> - ^dM^^''^

Z (t^) = E (e^") • • T(.) • exp{-2v(M)£)

S Z ('^) - Z (_) O 3.

(-1) • 1 ^

:1 -r z^U) eKp[-2y(m)i]

(3.21)

-27-
= E (e-^^) Z(i^) + Z (o)) o g

T(u)) • exp(-2Y(a))JlJ

1 + P_(w)

1 - p (w)p^(a)) exp[-2Y(a))£)J

Using the homomorphic deconvolution technique, each of the two equations, eqs. (3.21)

and (3.22), can be split into two components, the slowly varying component and the rapidly

varying one (subscripted by s and r in the following equations, respectively).

E_ (eJ-^) = E (eJ")

Ss g

Z (cj) + Z (u) o g

T(a)) • exp(-2Y(tj)£) • (-1)

1 + P (oo)

1 + P (w)-exp(-2Y (w)«.]

(e-J ) = E (e-J

) Ls B Z (w) + Z (w) O 2

T(a)) • exp(-2Y(a)))l) • p^(a))

1 + P_(a))

1 - p (w)'Pj^(u))'exp(-2Y(a))X.)

The above four equations together with eq. (3.14) can be used to solve for p (w), p (oi)

,

iw L g

y(oj), and the product Eg(e )T(a))

The homomorphic deconvolution technique applied to TDR problems not only gives access

to more of the microwave network parameters but also works for overlapping reflection

cases, where the conventional methods fails.

3.4.2.4 Application

A computer program was written to simulate the TDR network of figure 3.43b with parallel

R-C combinations to serve as reflective load and generator impedances. The homomorphic

deconvolution technique was applied to the waveforms of the simulated model; the network

parameters and various waveforms are given in figures 3.45 through 3.50. In the numerical computations for the present example 1024 samples were used. The

time and frequency sampling intervals were 2.45 ps and 0.4 GHz, respectively.

Equations (3.23) and (3.25) were used to calculate the reflection coefficient Pj^(w)

from the transform domain data of the separated signals (shown in the time domain, figures 3.48

and 3.50), Then the resultant Pj^(<^) was used to determine R and C of the load impedance

-28-
Z (o)) . The calculated values of R and C vs. frequency, figures 3.51 and 3.52, oscillate

Li around their respective actual values. These erroneous oscillations which correspond to those indicated in figures 3.48 and 3.50 are again due to (a) the finite number of samples

considered, (b) the finite observation time window, and (c) the imperfect separation in the

cepstrum due to overlapping long tails of the different components.

3.5 Summary and Conclusions

The present chapter presented several classes of time domain measurement problems in

the field of linear microwave network analysis. The homomorphic deconvolution transform,

that was developed in chapter 2, was used in deconvo luting various convolutions in these

problems. The examples that were considered are as follows:

(1) Pure resistive loading along a transmission line.

(2) Commensurate transmission line networks.

(3) TDT type measurement with two reactive discontinuities.

(4) Conventional TDR measurement to determine a reactive load impedance.

The homomorphic deconvolution method was proven to be very effective in separating

convolutions. The method was not limited by the overlap of consequent reflections in the

time domain.

-29-
SAMPLING

OSCILLOSCOPE

MICROWAVE

NETWORK

a. TDR Set Up.

MICROWAVE

NETWORK

SAMPLING

OSCILLOSCOPE

b. TDT Set Up.

gure 3.1. Time Domain Measurements

-30-
-31-
w • CnC

•H •rH 'd0 C

0H

W

> 0) •H •H P e

tn 0) •H

cn fO

Q) >-i t^

QJ (T3 C

0

cn

Q)U

:3 •H

Cm
I c

>-

>-

o

cn

-H

co

•H

U}

(0 •He

COc

Eh

Q) +J

to

5-1 :3COc

0)

o

u

-H

Cm

-33-
a. NETWORK

eg(t)l

TIME

0

2t,

SIGNAL LEVEL

1

2 2 2t^+4t2 P1P2 (l+P-])

2t^+6t2 P^^P2"^(1+P] )^

2t^+8t2 P^"^P2^(1+P]

c. IMPULSE RESPONSE

1 .0

.9

•8

.7

.6

.5

.4

.3

.2

.1 h(t) 0

-.1

. -.2

-.3

-.4

-.5

h(t) = 6(t)+h'(t)

.2, p2

= -.75

1

' '2 WHERE I =84.67 ps

1 wv • — oY

0

1 .

l-<

<

^1 >• -< >

3t 5t 7t 9t Tlx

-.2

072

-.0108 .00162

.48

Figure 3.5. Case of Two Reflections.

-34-
-35-
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CON

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-37-
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-45-
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NETWORK

i

MULTIPLE REFLECTION PATTERN

TIME SIGNAL LEVEL

0

2t.

2x^-2x2

2x^.4x2

2x-| +0X2

2x^-3x2

INPUT RESPONSE

1 .0

.9

.8

.7

.6

.5

.4

.3

.2

.1

0

-.2

-.3

-.4

-.5

h(t) = 6(t)+h'{t)

p

= 1/3

= 1.5t,

X2 = 1 .2^

WHERE - = 84.6 ps

3333

'2

5.4x

3x

7.8x

1 -.0329

10. 2x

-.00366

2963

12.6x

.000406

Figure 3.20. Case of Two Commensurate Transmission

Lines with Two Discontinuities.
-50-
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-54-
-55-
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-75-
Figure 3.47. (t) = e^^. (t) - e^^(t)

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Figure 3.48. e_ (t) , Arrows Indicate Error Oscillations

LS

-76-
Figure 3.50. (t) , Arrows Indicate Error Oscillations.
100
4. TIME DOMAIN DIELECTRIC SPECTROSCOPY

4.1 Introduction

In chapter 3 the homomorphic deconvolution transform was successful in solving various

time domain network analysis problems, and more specifically, time domain ref lectometry and

transmission problems. Network parameters of interest were identified, and more importantly,

the time domain overlap of multiple reflections did not put any limitations or restrictions

on the method.

In the present chapter another application of time domain measurements will be considered. Here, the homomorphic deconvolution transform will be applied to time domain

dielectric spectroscopy (TDDS); i.e., the study of dielectric relaxation phenomenon by time

domain methods.

TDDS is in fact a TDR problem, but with the emphasis being turned to the dielectric

material filling the transmission line in order to evaluate the frequency dependent complex

dielectric constant (relative permittivity) k(w)

As was indicated in chapter 3, the conventional TDR method is based upon and limited

by time windowing to select the relevant or desired waveform epoch. Thus, for the con- ventional TDR method to be used, the dielectric sample has to be either relatively infinitesimal or relatively infinite in length to obtain nonoverlapping time windows. The

homomorphic deconvolution method will be applied in the present chapter to remove these

restrictions and allow the use of finite size samples.

The method will be applied to a model typical of dielectric materials having polarization absorptions over the frequency range from 1 kHz upward to 100 GHz.

The typical experimental setup for a TDDS measurement [12-18] is shown in figure 4.1.

The dielectric material to be characterized is formed into the shape required to fill a coaxial sample holder of finite length d corresponding to an empty precision line having a characteristic impedance Z^(w) and a propagation constant y(<^)' A time domain waveform

e^(t) is to be recorded. This waveform consists of the multiple reflections from both

dielectric-air interfaces as well as the incident wave. Because the physical dimensions of the two transmission lines are the same, the characteristic impedance Z^(a)) and the complex propagation constant Yj(w) of the dielectric

filled transmission line are related to the corresponding quantities of the uncelled air

line as follows:

4.2 Theory of Time Domain Dielectric Spectroscopy

Z (w)/A:(w)

o (4.1)

and

Y (u) • vk(oo) o (4.2)

-79-
where k(a)) is the complex dielectric constant of the sample. The reflection coefficient at the air-dielectric interface on the line p (w) is given by

p(oo) = [Z^(a)) - Z (w)]/[Z,(ai) + Z (o))] o a o (4.3)

which upon substituting eq. (4.1) takes the form

pM = [l-/k(w)]/[H-/k(a))]. (4.4)

The form of eq. (4.4) can be solved for k(oj) as follows:

A(a3) = (l-p(a)))/(l+p(cj))

br

k(a)) = [(l-p(a)))/[l+p(oo))]2. (4.5)

This equation, eq. (4.5), summarizes the concept of using the TDR method in dielectric

characteristics by experimentally observing the reflected wave produced by the reflection

coefficient p (w) . The deduced reflection coefficient p(a)) is then substituted in eq. (4.5)

to yield the complex dielectric constant k(a)). In the conventional TDDS method [12-18] the first reflection coming back from the airdielectric interface determines the measurement's time window; i.e., the reflected waveform

is observed for the time interval containing the first reflection. This first reflection

has a Fourier transform which is the product of the Fourier transforms of the incident

signal e (t)/2 and the reflection coefficient. The separability of the time window and the

knowledge of ^g(t) allow the calibration of p(a)) and ultimately k(u)

.

As mentioned in the introduction section, the separability of the first reflection

requires either an infinitesimal or an infinite sample length. The requirement of having a very long sample is not generally practical. On the other hand an infinitesimal sample

would not alter ^g('^) ^° great degree, thereby not imparting to the signal waveform

much information about the dielectric. A finite sample of practical size will produce

overlapping multiple reflections thus invalidating the conventional (windowing) method. In such cases, the homomorphic deconvolution transform will succeed in removing (deconvoluting) the effects of multiple reflections allowing the determination of p(co).

The frequency domain form E (e-''^) of the recorded waveform e (t) is given by

By virtue of eqs. (4.2) and (4.4), the above eq. (4.6) has only one unknown k(a)) provided

that Ego(e-''^), Z (w), and y M are known. However, such an equation can not be solved, as

o it is, to yield this unknown. Some research workers [12] have gone through approximations

to simplify this equation; but the use of the homomorphic deconvolution transform on

eq. (4.6), as it is, will yield the required information needless of any assumptions or approximations.

1 - exp (-2Yj (w)d)

(4.6)

-80-
4.3 The Homomorphic Deconvolution as Applied to TDDS Problems

The homomorphic deconvolution transform that was first developed in chapter 2, and

then used with TDR networks in chapter 3, separates a convolution into its two components.

These two components are the slowly and rapidly varying frequency domain components. By

examining eq. (4.6) the slowly and rapidly varying components can be identified as

E (e>)

(e^") =-^-5 exv[-2y(ui)l] p (oo) , (4.7) Is / o

and

1 - exp f-2Y , (co)dl = (4.8)

^ 1 - p^(a)) exp (-2Y^(tD)d)

respectively. In these two equations, eq. (4.7) is a simple one that can be solved in

p(u)), and consequently k.(to) , while eq. (4.8) still has the same complications that originally

existed in eq. (4.6).

4.4 Model for Debye Dielectric [9, 11]

One well known class of dielectric materials will be considered in the application

presented in this chapter, the single relaxation time dielectric known as the Debye dielectric. The complex dielectric constant k(a)) of a Debye dielectric is given by

^ ~ K

k(w) = + — (4.9)

1 + jtOT

where k = k(to=0) , k = k(u='») , and t is the dielectric relaxation time. This equation is

O CO commonly put in the form

k(w) = k'(uj) - jk"(ca) (4.10)

where

k'(aj) = (k^+uj2^2i^^-)/d+^2^2) (4.11)

and

k"(u) = wT(k -k )/(l-Ko2^2)_ (4.12)

O CO The plot k"(w) vs. k'(u) is a semicircle of radius r and centered at the point (c,0)

where

r = (k^-kJ/2 (4.13)

and

^ = (V^-)/2. (4.14)

the point (c,r) is the point at which to = u^, where

V = -81- (^-15)
The semicircle is shown in figure 4.2; this semicircle is commonly called the Debye semicircle or the Cole-Cole plot [10-11].

It is important to mention that the complex dielectric constant parameters k^, k^, and t can be determined once the dielectric semicircle is obtained. This is achieved by virtue

of the eqs. (4.13), (4.14), and (4.15) to be

k = c + r, o

k = c - r,

00

and

T = l/oj . o

Also the circular geometry of the graph provides a means of consistently smoothing experimental data for k^(a)) and k'"(aj).

Finally, the point should be made that the TDDS methods developed here and not restricted

to any particular dielectric model. The Debye model has been chosen because it is a well

known model for the dilute mixture of polar molecules in nonpolar solvents. As such the

model will clearly demonstrate the utility of the deconvolution method in obtaining the

dielectric parameters, k''(aj) and k"(aj).

In general, a dielectric material may contain a distribution of relaxation times

comprising its relaxation process. The relative dielectric constant may be viewed as a sum

of relaxation processes each having a distribution of relaxation times [19].

oo , k(a)) = k + / dT

o

where g(T) is the relaxation time distribution function.

4.5 Application of Homomorphic Deconvolution in TDDS [20]

A computer program was written to simulate the TDDS setup of figure 4.1. The transmission line characteristic impedance was 50 U and the dielectric sample length was one cm. The dielectric parameters were chosen to by typical of actual dielectrics satisfying the

Debye model. The relative dielectric constant employed was

^.r ^ / « J. 24.5 - 4.8

k(a)) = 4.8 +

1 + j2.0xl0 ^^w

in which k =24.5, k =4.8, and t = 2 ps. The time domain waveform of the generator

o =° e (t) is the same one used in section 3.4.2 of chapter 3; it is given in figure 3.46.

S -1 The homomorphic deconvolution transform was applied to the computed signal E^(e-^ ) to separate both its slowly and rapidly varying components. The slowly varying component

together with eq. (4.7) were used to calculate the reflection coefficient p(a)). The resultant values of p(a)) were substituted in eq. (4.5) to yield the desired k(uj) . Figures 4.4,

4.5, and 4.6 show the results of the above mentioned steps. Figure 4.4 gives the magnitude

of E^(e-'*^); figure 4.5 gives the magnitude of the separated slowly varying component while

-82-
figure 4.6 shows the Cole-Cole plot of the calculated complex dielectric constant components

(Argand diagram). A semicircle was fitted [21] to the computed plot and also is given in

the figure. The cemicircle parameters were r = 9.80255, c = 14.60045, and = 0.50071 x

10''"^ rad/sec.

Upon substituting these values in eqs. (4.16), (4.17), and (4.18), the dielectric

parameters are evaluated to be

k =24.403,

o

k = 4.7979, CO

and

T = 1.997 ps,

which are within 0.3% error of their originally given values.

4.6 Summary and Conclusions

In this chapter, the theory of the time domain dielectric spectroscopy measurements

was presented. Conventionally this method of measurement works only for non-overlapping

reflections; i.e., reflections occurring in separate time domain windows. If the reflections overlap, only approximate solutions may be applied to the frequency domain form of

the signal.

By applying the homomorphic deconvolution transformation to the TDDS waveform, a new method was developed which was not limited by overlapping reflections. The new method was applied to the well known Debye dielectric model. The method eliminated the overlapping

reflections and determined the dielectric parameters to within 0.3% error of their model

values

The new method is a general one and has a great utility for time domain measurements

on dielectrics since it does not include any restrictions on the dielectric material

parameters or require the assumption of a prior physical model. Furthermore, the new method is not restricted to time domain spectroscopy of only dielectric materials but has

utility for time domain spectroscopy in general such as for magnetic and other phenomena.

-83-
-84-
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-89-
5. THE APPLICATION OF THE HOMOMORPHIC DECONVOLUTION TRANSFORM TO THE REMOVAL OF SCATTERING COMPONENTS FROM ANTENNA TEST RANGE DATA

5.1 Introduction

Time domain analysis of electromagnetic radiation problems has become increasingly more popular in the recent years because the time domain method provides physical insight

into the behaviour of antenna structures excited by pulsed sources. Time domain measurements do offer significant advantages over the classical steady-state frequency domain

measurement; they are as follows:

a. There is no need for an anechoic chamber to absorb unwanted reflections because

time-windowing can be shut-out reflections.

b. There is no need for repeated magnitude and phase measurements at different frequencies because the time domain waveform contains the complex spectral information in

a single entity.

Frequency domain information can be obtained from the measured time domain waveform by

simply computing the corresponding Fourier transform. However, a recently applied method

[22, 23 and 24] based upon Prony's algorithm [25] can also be used to deduce the required

frequency domain information. The method is based on the fact that the time-dependent

function can be expressed as the sum of complex exponentials

a.t • f(t) = ^ A. e

^ (5.1)

i

for which the frequency domain form is

A.

v-^ ' ^

. \

; , where are the complex poles of F(s) and A^ are the corresponding complex amplitudes or residues. By fitting the time domain data to the form of eq . (5.1) and determining the

location of the singularities a^'s and the corresponding residues A^'s, the system under

measurement is fully characterized according to classical circuit theory.

This method has the advantage of yielding an analytical spectral response for the

experimental system rather than the discrete form that the numerically implemented Fourier

transform yields. Also, it is capable of extrapolating the measured transient response for

both early and late times.

It is clear that the above discussion favors time domain measurements for pulsed

electromagnetic radiating systems. Also, it is clear that it should be possible to windowout any unwanted reflections and scattering components. However, for cases where these

undesired reflections or scattering components do overlap on the useful part of the time

domain signal, the homomorphic deconvolution should enable the separation and the rejection

of the unwanted components.

-90-
The present chapter will demonstrate the power of the homomorphic deconvolution transform in separating and removing unwanted reflections and scattering components in a pulsed

electromagnetic radiating system. The transform will be applied to radiation from a wideband horn antenna pulsed by baseband signal having frequency components out to beyond

6 GHz.

5.2 The Homomorphic Deconvolution as Applied to Antenna System

The homomorphic deconvolution transform was developed earlier in chapter 2, and was used in chapters 3 and 4 for the separation of multiple reflections due to different discontinuities on a transmission line. In these cases point or discrete discontinuities

generated reflections which transformed to clusters (discrete well-separated pulses) in the

cepstrum-domain. The localized clusters were then separated by filtering.

In radiation systems, the situation is different. In addition to the point reflections

there exist plane reflections, which can be called a continuium of multipaths. Also, there

exist scattering components due to the interfaces between the antenna aperture and the

radiation medium, and also those due to different nearby scattering objects. Figure 5.1

illustrates some of these phenomena.

Consequently, it is not expected that definite or point located clusters will appear

in the cepstrum-domain of such systems; rather, it is expected that a distribution of

contributions will appear in the cepstrum-domain. Figure 5.2 shows a typical cepstrum for

antenna systems.* This, of course, makes the separation of unwanted components more difficult than the cases of chapters 3 and 4. Accordingly, the degree of filtering must be

determined by a trial and error procedure. If too much of the cepstrum spread components

is removed, it may include parts of the useful signal information. On the other hand, not

removing enough of the unwanted components may result in a remainder which overlaps the

useful time domain signal.

The separation in the cesptrum domain then should be done on a successive trial basis

using different filters; the results should be examined in the light of the physical system

to decide upon the best filtering process.

The next section of the present chapter will be devoted to the description of the

antenna system to be used in the forthcoming application.

5.3 The Antenna System

The experimental setup that was used is shown in figure 5.3. It consists of two wideband antenna horns set as a transmitting receiving pair, an impulse generator feeding

the transmitting horn, and a wideband sampling oscilloscope connected at the output port of

the receiving antenna to display the received waveforms. A layer of absorbing material was placed such that it would prevent reflections from the floor. When reflections were

desired, the absorbing material was replaced by a large metallic sheet.

*Figure 5.2 may be compared with figure 3.10 to show cepstrum spread as compared to clusters

-91-
The antenna horns [26] are identical in their physical structure which is shown in

figure 5.4. The horn is designed in a symmetrical fashion around a center ground plane.

The antenna presents a 50 ohm driving point impedance which minimizes reflections in the 50

ohm transmission lines associated with the transmitter (impulse generator) and receiver

(oscilloscope). , The impulse generator has a 50 ohm source impedance and generates the pulse shown in

figure 5.5. The oscilloscope sampling head terminates the receiving transmission line in

50 ohm.

~

'

' "- ' • 5.4 The Application

The output signal of the receiving horn of the setup shown in figure 5.3, with the

absorbing material in place, was acquired and is shown in figure 5.6. The figure shows some reflection and scattering components not overlapping the main signal. Therefore,

ordinary time windowing can be used to remove these unwanted components. However, the

homomorphic deconvolution transform will be applied for the purpose of demonstrating the

effect of the separating filter on the separated output signal.

Applying the homomorphic deconvolution transform on this signal, the cepstrum of

figure 5.7 is obtained, which, as was discussed earlier, contains a spread of contributions

rather than discrete clusters. The separation process was tried twice on this cepstrum

•using different filters having the half-widths A and B, respectively, as indicated in

figure 5.7 (the filters were symmetric about the cepstrum origin). The result of the first

separation at position A is given in figure 5.8, which may be compared with figure 5.9

where the reflections were time-windowed out at the indicated point on the figure. The two results very closely match, while the separation at B, shown in figure 5.10, matches the

time-windowed signal of figure 5.11. From these figures it can be concluded that the narrower the window in the cepstrum the narrower the window in the corresponding time

domain. Also, a too narrow window in the cepstrum will start distorting the main signal as is clear from separation B where some distortion showed up just preceding the main signal,

figure 5.10b.

After experimenting with the above separation trials, it is interesting now to try a case where the reflections overlap the main signal. This was achieved by introducing a large metallic plate in place of the absorbing material in the setup of figure 5.3. The

observed waveform is given in figure 5.12 where it is possible to see the reflections from

the metallic plate overlapping the main signal. The corresponding cepstrum, obtained by

the application of the homomorphic transform, is shown in figure 5.13. Two separations were used again at the windows indicated in the figure. The results of the two separations

are given in figures 5.14 and 5.15. The two separation results are slightly different from

each other and from the clean signal of either figure 5.9 or 5.11, but it is important to notice that the introduced overlapping reflections (by the metallic plate) were successfully

removed through the use of the homomorphic deconvolution transform.

-92-
5.5 Summary and Conclusions

The present chapter began with an introduction about the role that the time domain

measurement techniques can play in solving electromagnetic radiation problems. Then the

distributed nature of the expected cepstrum in such problems was discussed and compared

with previously studied discrete cases . Two examples of applying the homomorphic deconvolution transform to an antenna system were presented, and the transform proved to be very

successful in the removal of the unwanted reflections and scattering components either

overlapping or not overlapping the main signal containing the desired signal information.

-93-
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-95-
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DIELECTRIC FILLING

Figure 5.4. The Wideband Antenna Horn.

(all dimension in cm)

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-113-
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-117-
6. TRANSFER NETWORK IDENTIFICATION (MODELING) OF FEED-THROUGH SAMPLING-HEAD

- 6.1 Introduction

In chapters 3 and 4 it was noted that a given oscilloscope recording a time domain

signal always presents a recorded waveform different from the signal waveform, which is due

to the limited oscilloscope bandwidth. The recorded signal is the convolution of the signal

waveform with that of the oscilloscope impulse response, figure 6.1. Also, it was shown in

these chapters that the oscilloscope bandwidth limitation did not prevent the application of

the homomorphic transformation to solve problems in which the oscilloscope transfer function

was eliminated by insertion (ratio) measurements. However, if it is necessary to determine

the true waveform at the input of the oscilloscope, then a model for the transfer properties

of the oscilloscope is required.

In time domain spectroscopy, the overall bandwidth is determined by the product of the

bandwidths of the signal source and the oscilloscope. In order to analyze the relative

bandwidth contributions, it is necessary to determine a model for the oscilloscope.

For the picosecond time domain signals which are being dealt with in this work, the

wideband sampling oscilloscope is used for observing and recording such signals. The one

kind of sampling head that is most commonly used in time domain ref lectometry measurements

is the feed-through sampling-head because of the convenience of usage in observing incident

and reflected time domain signals on the signal line, figure 6.2.

The aim of this chapter is to develop a network model based upon the physical properties

of the sampling-head. Nominally, the manufacturer claims a 28 picosecond (10% to 90%)

transition time. The network model is to be used to predict both the oscilloscope's time

and frequency domain responses. The predicted model responses are to be used to deconvolve

the observed signals in order to deduce the true waveform at the oscilloscope input port.

Previous workers have attempted to use heuristically based mathematical models which

considered only an idealized sampling process [27].

6.2 Theory and Construction of Sampler

Previous publications [28,29] went in depth into the theory and construction of the

feed-through samplers; only a brief coverage will be presented here.

Figure 6.3 shows the balanced two-diode sampling circuit; this circuit is the basic

structure in the feed-through sampler. The circuit works in the following manner. When

the signal on the feed-through line is zero, application of a balanced strobing (sampling

command) pulse on the two diodes will result in equal conduction in the diodes; consequently,

an equal charge storage results on the sampling capacitors C^ which in turn yields a zero output. But if the signal on the line has a nonzero value, nonequal diode conduction will

result in a nonequal charge storage on the sampling capacitors and an output different from

zero. The output polarity and magnitude are related to the input polarity and magnitude.

The function of the feedback is to readjust the diodes' reverse bias after each sampling

such that the next sampling process only detects the change (the difference) in the input

-118-
signal. The feedback serves two purposes: (1) it increases the dynamic range for the

allowable input voltage to the sampler, and (2) it maintains the operating region of the

diode almost constant during each successive sample.

Since the output of the sampling circuit is a measure of the differences or level

changes in the input signal, an integrating circuit is needed to give the required measure

of the input signal. Figure 6.4 shows the dynamics of the sampling network assuming a ramp

input signal and a triangular sampling pulse.

The detailed view of the physical realization of the two-diode feed-through sampler is

shown in figure 6.5. In the figure it is seen that the feed-through transmission line is

tapered or stepped-down on both the input and output sides; this is successively done through

a three section compensated stepped-taper maintaining a 50 ohm impedance along the line

(details shown in figure 6.8). The sampling diodes are put in a perpendicular fashion to the transmission line touching the inner conductor from both sides . In this region the

inner conductor is a thin curved piece of metal that joins the two tapered sections together.

The outer conductor forms a biconical cavity, which acts as a shorted biconical transmission

line which shapes the strobe pulse, as shown in figure 6.6. The biconical cavity is dielectric filled to mask the packaging capacitance of the sampling diodes to lessen the effect of

the diode on the 50 ohm characteristic impedance throughout the cavity.

The sampling capacitors, 2 pf each, are built in the sampling diodes; figure 6.7 shows

the details of a diode pack.

6.3 Modeling of Physical Structure

Precise measurements were performed on the physical dimensions of a disassembled

sampling-head; the results are given in figure 6.8. Each of the step line discontinuities

can be modeled by an equivalent capacitor [30,31]; the value of this capacitance is a function of the inner and outer diameters of the discontinuity. The discontinuity dimensions

and capacitances are tabulated in Table 6.1. Figure 6.9 shows the equivalent circuit for

the tapered structures complete with the characteristic impedances of line portions in

between.

6.4 Biconical Cavity

The details of the cavity have been shown in figures 6.5, 6.6, and 6.8. The cavity

parameters of interest are (1) the characteristic impedance as seen by the signal along the

feed-through line, (2) the characteristic impedance of the biconical transmission line as seen by the strobing pulse, (3) the travel time out and back along the shorted biconical

line which determines the shape and duration of the sampling pulse applied to the two-diode

sampling gate, figure 6.6, and (4) the characteristic impedance of the wire line carrying

the strobing pulse down to the cavity.

-119-
Table 6.1.

Discontinuity Dimensions (mm) Capacitances (pf)

a

" b c d C m. C

out

6 . 9977 3 .0429 4. 9835 2 . 1636 .00607 .00884

4. 9835 2 . 1636 2 . 8575 1 . 2413 .00803 .01016

2. 8575 1 .2412 2. 0320 0 .8890 .00246 .00349

2. 0320 0 .8890 0 .6604 .00170

2. 0447 0 .8814 0 .6604 .00170

2. 8448 1 . 2446 2. 0447 0 .8814 .00246 .00349

4. 9962 2 .1654 2. 8448 1 . 2446 .00803 .01016

7 . 0104 3 .0218 4. 9962 2 . 1654 .00607 .00884

7 . 0104 3 .0218 2 .0244 .00780

7 . 0104 3 .0231 2 .0244 .00780

-120-
To find (1), (2), and (3) above, two TDR measurements were performed on the cavity with

and without the dielectric filling, with the diodes removed in both cases. The results are shown in figure 6.10. Figure 6.10a clearly shows that the feed-through line is within

± 1 ohm from the 50 ohm characteristic impedance; i.e., the reflected voltage is no greater

than ± 5 millivolts. Also, it is seen that there is an inductive effect in the central

portion of the cavity due to the metallic curved strip that connects the tips of the two inner conductor tapered sections. From figure 6.10b, it is found that the absence of the

dielectric filling caused a change in the impedance level that corresponds to a reflection

coefficient of p = 0.196, but since p = (v^-1) / (/k+1) , where k is the dielectric constant, k

is obtained from = (l+p)/(l-p) = 1.49 which gives

k = 2.21. (6.1)

The value of the dielectric constant k is to be used to calculate the travel time , be

figure 6.6. From

= ^

d^ /c (6.2)

be be

d^^ is the corresponding travel distance which is equal to 4.48 mm as given in figure 6.8,

and c is the speed of light; therefore

1

= 1.49 X 4.48/(3x10 ) = 22.2 ps. (6.3)

DC

Tne third cavity parameter to be determined is the characteristic impedance Z^^ of the

"shorted" biconical line which can be calculated using the relation [32],

Z^^ = (l20 in cot (a/2) J /vie ohms (6.4)

where i is the bicone half angle which is equal to 38° 22' as given in figure 6.8. The

computed value of Z^^ is

Z^^ = 85.2 ohms. (6.5)

Turning now to the wire line which carries the strobing pulse to the biconical cavity,

the characteristic impedance Z of the wire line is to be estimated. The inner conductor of

^ w

the line has an outer diameter r^ = 0.127 mm, coated with varnish to a diameter = 0.33 mm, while the inner diameter of the outside conductor is r^ = 0.432 mm. An estimate for the

value of the dielectric constant k of the used varnish is k =2. The impedance Z can V V w

be written as the sum of the two impedances Z , and Z ^. ^ wl w2

Z = Z , + Z - = —- In — +60 in — W wl W2 /r- /k r^ r„ V 1 2 60 „ 0 .33 ^ . 0.43

= — In + 60 j^n /2 0.127 0.33

= 57 ohms. (6.6)
6.5 Sampling Diodes

The sampling diodes that are being used In the feed-through sampler are hot carrier

(Schottky) diodes; the electrical model of such diodes [33,34,35] Is shovm In figure 6.11,

in which ,.. , ; ,

-

.

I

s R.

J C.

2 R

s ^d

C

P

the diode (reverse) saturation current,

the junction's dynamic resistance,

the junction capacitance,

the spreading and contact resistance,

the lead inductance, and

the package capacitance,

wnere

R. = 1/

31.

'•3V, (6.7)

and

J exp(v./(f^)l - 1

:fl-.. (6.8)

In the following, the measurements required for the evaluation of the above diode

parameters are divided into two types: (1) static to evaluate and and (2) dynamic to evaluate C , L , , and C..

P d J

6.5.1 Static measurements (I and R ) s s

The circuit of figure 6.12 was used to measure the diode's dc characteristics by taking

values of the dc current I, for various values of dc diode voltage V,; the measured values

Q d are plotted in figure 6.13. In this dc measurement, the diode currents and voltages are

^j =

^d' (6.9)

V_, - I.R . d J s (6.10)

In the forward bias case where I. >> I , the diode I-V characteristic, eq. (6.8), can be

J ^ approximated as

I. = I exp(V./<))) (6.11)

£n I. = Jin I + V./(j); I. >> I

J s J ^

J s (6.12)

where

(p = kT/e.

Substituting eq. (6.9) in eq. (6.12) gives

Jin I, = Jin I + (V,-I,R )/(}); I, >> I . d s dds d s (6.13)

-122-
The experimental data of figure 6.13 were fitted to eq. (6.13) above yielding the following

values for R and I : s s

= 19 ohms (6.14)

= 2.76 X 10 amps. (6.15)

6.5.2 Dynamic measurements (C , L,, and C.)

^ p d J

To determine the sampling diodes' dynamic parameters (C^, L^, and C_. )

, the TDR technique was used; the setup for the experiment is shown in figure 6.14. The test signal was

the tunnel diode pulse given in figure 6.15. Because of the limited resolution due to the nonzero risetime of the test signal, the performed TDR measurements were not able to directly

resolve the contributions of the sampling diodes from those of the central conductor inductanc

L^; i.e., the TDR views the sampling diodes and the central inductance as if they were at the same physical location. This lumping can be tolerated (and later resolved) since the

metallic strip extends for about 1 mm on both sides of the diodes in a symmetrical fashion.

This symmetry allows L to be divided equally into two equal parts, and L„ on the left

C L K

and right of the diodes, respectively, figure 6.16.

To overcome the TDR lumping of and the diodes' contributions, two TDR measurements

were performed; one with both diodes in place, and the other one with only one of the diodes

in place, figures 6.17a and 6.17b. Denoting the maximum voltage drop-peaks as e^ and

respectively, the TDR voltages are ^ = 2e^ + (6.16)

62 = + (6.17)

where e^ and e^ are the contributions in the TDR signal corresponding to and diode,

respectively. Solving eqs. (6.16) and (6.17) for e and e, yields

Li Q

e^^ = 2e2 - e^ (6.18)

= - e^. (6.19)

Substituting for e^ and e^ from figure 6.17 gives

e^ = -9.9 mV (drop-peak) = 9.9 mV peak (6.20)

e^ = 19.7 mV. (6.21)

The value of eq. (6.20) can be used to evaluate the inductance using the relation [36]

2Z L = — e- (6.22) m L ^ '

where m is the maximum slope of the test pulse, figure 6.15. Therefore,

= ^^^-^ ^ 9.9 X 10"^ = 150 ph (6.23)

-123-
and

L^= 75 ph. (6.24)

To measure the package capacitance of the diode, a third TDR measurement, figure

6.17c, was done with one working diode and an open diode; the open diode contributes only

Denoting the voltage drop-peak in this measurement e^, the TDR voltage relation is

^3 = ^d + % + . (6.25)

where e is the contribution of the open diode, or in other words, the C contribution.

P P Consequently, ., % = ^3 ^d " ' (6.26)

Upon substituting for e^ from figure 6.17c, and for e^^ and e^ using eqs. (6.20) and (6.21),

the value of e is given by

P

e = 20.8 - 19.7 + 9.9 = 11 mV. (6.27)

can then be calculated using [36] •n

2_

"o

which yields

2

•-- - , C = X 11 = .067 pf . (6.29)

P 6.58 X 50

For the order of magnitudes involved in the present TDR measurement, computations

showed that the effect of diode inductance on the voltage drop-peaks is minor; i.e.,

such measurements are not able to yield the value L^. However, previous measurements

[37] that were done on the same particular sampler gave a complex value 14 + j5 ohms at

10 Griz for the impedance Z^/2 of the sampling gate with a diode biasing forward current of

5 ma. This information can be used to calculate the value of L, as follows: the measured

d

impedance at the sampling gate is in fact the impedance of the parallel combination of the two sampling diodes, each one in series with its sampling capacitor C^; this combination is

shown in figure 6.18. At 5 ma diode forward current, the junction resistance is

R. = 1/

J

81.

3V.

J

<^/l^ = 5.2 n (6.30)

R./2 = 2.6 n. (6.31)

J

Since C. is expected to be lower than 0.1 pf, the reactance of 2C^ at 10 GHz is higher than

80 which is much greater than the reactance of 2C^ can be neglected in this

-124-
particular case. Hence, the impedances can be expressed as

Z (l/ja)2C )

• (R /2+R./2+ja)L,/2) _s. 1_ + 2 § a 3 . (6,32)

2 jw2C (l/j(jj2C ) + (R /2+R./2+ja)L,/2)

S P SJ Q

Since C , R , and R. are known, eq. (6.32) above can be solved to give

p s' J

= 300 ph. (6.33)

Now, the only component that is left to be determined is ; and since it is known to be a function of the junction voltage, a set of TDR measurements, with both diodes in place, was performed at different biasing voltages. The biasing circuit was that shown in figure

6.19, and in figure 6.20 only two of the various TDR waveforms are shown; however, all the

results are listed in table 6.2. Denoting the voltage drop-peak in these TDR runs by e(v),

where v is the diode bias, the TDR voltage is

e(v) = 2e + 2e.(v) + e^ (6.34)

P J ^

from which e.(v), the contribution due to C . , can be obtained as

J J e^(v) = [e(v) - - 2e^]/2. (6.35)

The corresponding (v) is

C.(v) = -f- e (v). (6.36)

These steps are shown also in table 6.2, and the calculated C.(v) is plotted in figure 6.21.

By evaluating (v) , this phase of modeling the sampling diodes is concluded.

In summary the obtained values for the diode model, figure 6.11, are

C = 0.067 pf

P

= 300 ph

R = 19 n

s

Ig = 2.76 X lo"-*--"- amp

(v) is given in figure 6.21. (6.37)

Also, the value of the central conductor inductances has been determined as

L- = L„ = 75 ph. (6.38)

6.6 The Sampling Pulse

The experimental setup of figure 6.22 was used to observe the strobe pulse waveform;

only the leading edge of the pulse will be considered since it is the only part of the

waveform that forms the sampling pulse, figure 6.6. The observed signal is shown in

figure 6.23 which shows an amplitude of 9.58 volts and a 10% to 90% risetime of 53.5 ps;

both of these values have to be corrected as explained in the following discussion.

-125-
Table 6.2.

V V e(v) mV e , (v) mV

J C.(v) pf

3

0.0 26.86 7 .40 .045

-0.25 24 . 90 6 .42 .039

-0.50 23.58 5 .76 .035

-0.75 23.26 5.60 .034

-1.00 , 2 2.60 5.27 .032

-1.25 21.94 4.94 .030

-1.50 21.60 4.77 .029

-1.75 21.28 4. 61 .028

-2.00 20.62 4.28 .026

-2.25 20 .63 4.28 .026

-2.50 20.28 4.11 .025

-2.75 19.96 3. 95 . 024

-3.00 19.62 3.78 .023

-126-
The characteristic impedances, Z Z „, and Z „, of the stripline (on which the strobe

ol o2 o3

pulse was measured) , the (wire) microminiature coaxial line connecting the stripline to the

biconical cavity, and the biconical line are 50, 57, and 85.2 ohms, respectively, by virtue

of eqs. (6.3) and (6.4). These unmatched line impedances cause a voltage amplification

factor A given by

A = (l+p^)(l+P2) (6.39)

where the reflection coefficients and are determined as

^o2 ^ol 57 - 50 ,^ , =

z , + z , = 57T-50 = -o^^ ^^-"^^^ 02 ol

Z - Z 03 o2 85.2 - 57 „ mo l^\

'2 =

Z , + Z , = 85.2 + 57 = ^'^^^ ^^"^'^

o3 o2

from which A is calculated as

A = (1+.065) • (1+0.198) = 1.276. (6.42)

The amplitude of the observed strobe pulse vs. time has to be multiplied by the factor 1.2 76 to yield the corresponding amplitude at the input of the biconical shorted line.

The risetime correction has to take into consideration the attenuators, adaptors, and

measuring sampling-head responses. Experiments [38] have shown that elements have the

following nominal risetimes:

20 ps for 10 dB attenuator,

18 ps for 20 dB attenuator,

10 ps for each adaptor, and

20 ps for measuring sampling-head.

Using the risetime square rule for the cascaded setup of figure 6.22, the equivalent risetime T of this combination is nominally

e

T = /202 + 2 X io2 + 202 + 18^ = 36.4 ps. (6.43)

e

Using the square rule again, the estimated risetime of the strobe pulse is

X = /t2 _ x2 = /53.52 - 36. 42 = 39.2 ps, (6.44)

s m e

An assumed response having 36.4 ps risetime could be used to deconvolve the observed strobe

pulse to yield an estimate of the leading edge of the actual strobe pulse waveform. Another

method, which was found to be as good, is to scale the time variable in the observed waveform by the ratio t /i to yield an estimate for the actual waveform. The two methods gave

s m

values which were within 5% of each other.

The corrected leading edge waveform for the strobe pulse at the input port of the

shorted biconical line is given in figure 6.24.

-127-
: 6.7 Structure Step Response

In sections 6.2 through 6.6 all the sampling-head components were modeled. The com- plete model (equivalent circuit) would be the same as that shown in figure 6.9 except for

the cavity box to be replaced by its model, figure 6.25. To study the step response of the sampler, the computer program TDNA given in appendix C

is to be used; a voltage step is to be applied to the input port, and the sampled output is

to be integrated to yield the step response. As seen from figures 6.9 and 6.25, the number

of elements in the model is quite large; consequently, the amount of computer memory re- quired to consider this model was beyond the capacity of a large computer. The network

(model) of figures 6.9 and 6.25 was divided into some independent networks, in which case

the step response can be obtained in two steps: the structure step response and the sampling

process. Figure 6.26 shows the simplified network considered in the first step. Here the

short lines of figure 6.9 and the discontinuity capacitors on each side are replaced by one lumped capacitor, and two lines on both sides were matched in impedance to their neighboring

lines in impedance. In this way the number of lines and capacitors are reduced to about

one half. Also, in the figure the sampling signal and feedback voltage are ignored; the

diode bias is set to a constant negative value = -1.63 volts, at which the actual sampling

diodes normally operate, as was experimentally measured.

The network parameters shown in figure 6.26 were given to the computer as input data

for the program TDNA. The obtained time domain signal Vg at the node 8 is shown in figure

6.27.

6.8 Sampling Process

The second, and final, step in obtaining the sampling-head step response is to sample

the resultant voltage waveform of node 8 using the computer program SMPL given in appendix C. The voltage signal V„ is to be applied to node 17 in figure 6.28, each time the balanced

o strobing signal is applied to nodes 2 and 16. The reverse bias of the diodes is kept

at -1.63 volts, the normal operation value. The difference between the resultant voltage

increments on the sampling capacitors C , and C „ is a measure of the output voltage V

si s2 ^ ^ out

The sampling efficiency n [27,28] is defined as

m

As mentioned earlier in section 6.2, the feedback voltage readjusts the diodes' reverse

bias after each sample such that the next sampling process only detects the change in the

input signal; consequently, at the beginning of the sampling interval, the net diode bias

is a constant -1.63 volts. Accordingly, and referring to figure 6.28, the value of V

fb

for a new sampling process should be equal to V^^ of the previous sampling;

V^^ = V. . (6.46)

fb in

-128-
Figure 5.29 shows a block diagran of the sampling network and its feedback from the

figure, and using eqs . (6.45) and (6.46) it can be written that

fb out

= SG n V.m

= V. ; (6.47) in

therefore,

BGn = 1 (6.48)

or

-, V.

' BG = ^

= ^. (6.49)

out

To determine the factor 5G, a preliminary run for the computer program SMPL was done using

a small voltage step at node 17 as an input, and upon calculating its ratio to the output

voltage, the obtained values were

V. = .01 volt m

V = .0026907 volts

out

= .0026907 = (6-50)

out

The determination of the factor £G allows the calculation of the feedback voltage V„

ZD after each sample using eq . (6.47).

The program SMPL was run using the voltage Vg as the input signal to the sampling

network, and the input signal was sampled at 1 ps time intervals. The obtained output was normalized to yield the required sampling-head step response, figure 6.30. The step response

shows a 30.5 ps transition time.

Figures 6.31 through 5.34 show the voltage waveforms of the sampling process for one sampling interval. Figure 5.31 shows the time domain waveforzi of the resultant sampling

pulse ^2-± (voltage difference between nodes 3 and 1 in figure 6.28). The voltage across

one of the diodes ^ is shown in figure 6.32, while the voltage across one of the sampling

capacitors ^ is shown in figure 6.33. The difference between the two sampling capacitor

voltages ^2.2-1 ~ ^3-6 ''"^ sampled-data output, figure 5.3^. It is seen that the resultant

shape of the sampling pulse is somewhat different from being triangular, and the sampling

time (the time of diode conduction) is about 21 ps

.

In summary, it has been shown that the waveform of the sampling process or "sampling

pulse" is derived from the strobe pulse step like transition through shaping by the biconical

cavity into a quasi-triangular pulse which is in turn shaped by the nonlinear and bandlimiting properties of the sampling diode pair.

-129-
6.9 The Effect of the Diode Bias on the Sampling-Head Step Response

For ideal sampling, the sampling pulses should be infinitesimal in duration; a diracdelta function is the ideal shape. The actual sampling pulse, figure 6.31, does have nonzero duration; consequently, the step response of the sampling-head is expected to have

a nonideal (nonzero) transition time. From the theory of sampling processes it is clear

that the greater the duration of the sampling pulse, the slower will be the step response

transition time.

There is one way for the operator to control the sampling pulse duration and, consequently,

the step response; this is accomplished by controlling the diodes reverse bias, thus varying

the level at which the diodes conduct [39], figure 6.35. The larger the diode reverse

bias, the smaller will be the sampling time; consequently, the faster will be the response

of the sampling-head, and vice versa. However, there are limits to bias variations; very

low or very high reverse bias will cause imperfect sampling. Figure 6.36 shows the observed

waveforms of a tunnel diode step using the feed-through sampling-head for three values of

diode reverse bias corresponding to (a) bias for optimum sampling, (b) low reverse bias,

and (c) high reverse bias.

Additional tunnel diode waveforms were measured for various values of the diode reverse

bias. The 10%-90% transition time t was calculated for each waveform. The obtained t r r was plotted vs. reverse bias as in figure 6.37. Since the tunnel diode transition time is

constant, the observed variations are due to the sampling-head response variations. The

values of the sampling time for various values of reverse bias were computed (using the

program SMPL) and also plotted in figure 6.37. The figure agrees with the fact that the

smaller the sampling time the faster the response of the sampling-head, within the limits

of effective feedback control on the sampling level voltage. For reverse bias values

higher than 1.76 volts, the response is slower unlike the theoretical expectations. From

the figure it is clear that the fastest response for the sampling-head can be obtained by

adjusting the diode reverse bias to the value of 1.76 volts; the observed waveform at this

bias was slightly different from the normal sampling case (1.63 volts bias which results

from the manufacturer's alignment procedure for proper operation).

The sampling-head step response for the fastest response bias (1.76 volts) was com- puted and is given in figure 6.38, where it is seen that the step response shows a 27.5 ps transition time. Figure 6.39 gives the corresponding impulse response (time derivative).

6.10 The Deconvolution Routine

The final step in the present chapter is to develop a deconvolution routine that uses

the sampling-head impulse response given in figure 6.39 for deconvolving an observed i signal in order to deduce the true waveform at the sampling-head input port. In other

words, and referring to figure 6.1, the problem is to find x(t) knowing both y(t) and h(t).

The simplest deconvolution method known is to calculate the ratio of the frequency

domain transforms ¥(6"^^) /H(e~''^) to obtain X(e-''^), the frequency domain form of x(t) . This

method inherently produces a very noisy output, because in the division process the zeros

-130-
and the poles of H(e'^'^) must be exactly cancelled out. Any errors in the data representing

Y(e"''^) and H(e'''^) degrade the cancellation and introduce rapid variations (noise) in the

resultant X(e"''^) . In order to suppress the noise, low pass filtering and/or pole removal

processes have to be applied, the latter acting on poles introduced by incomplete pole/zero

cancellations. The pole removal processes are very sensitive which makes the use of the

simple ratio method for deconvolution rather unlikely.

Although more complicated deconvolution methods are available for use, they all involve

successive approximations to an estimated input x(t) to minimize the error between the

estimated output and the actual output y(t)

.

In this section a slightly modified homomorphic deconvolution method will be used

successfully to deconvolve the output y(t) to obtain x(t) . The method is advantageous in

its simplicity as well as yielding a low level of output noise in x(t)

.

6.10.1 The homomorphic deconvolution method

In the homomorphic deconvolution as was developed in chapter 2, the cepstrum of a convolution is the linear sum of the cesptras of the individual components of this con- volution. That is, using the notation of chapter 2 as applied to the signals of figure 6.1,

it can be written that

y(t) = x(t) + h(t) . (6.51)

Knowing both y(t) and h(t), the cepstras y(t) and h(t) can be used to determine x(t)

through the simple subtraction

x(t) = y(t) - h(t). (6.52)

The resultant x(t) can be processed (inverse-homomorphically transformed) back to the time

domain to yield the required x(t)

.

6.10.2 An application

To show the capability of the homomorphic deconvolution method as presented above, the

method will be applied to a known convolution. This known convolution y(t) was formed by

convolving the sampling-head impulse response h(t), figure 6.39, and an assumed typical

step-like time domain signal x^(t), figure 6.40. The resultant convolution y(t) is shown

in figure 6.41. The cesptras y(t) and h(t) given in figures 6.42 and 6.43, respectively,

were obtained by applying the homomorphic transformation to their corresponding time domain

forms. The difference x^(t) = y(t) - h(t) is shown in figure 6.44. Applying the inverse

homomorphic transform to x^(t) yields x^(t), the deconvolved estimate for the assumed

input signal x (t). Comparison of figures 6.45 and 6.40 shows that x, (t) matches to a

3i Q

great extent the original signal x (t) except for a very small amount of error oscillations

3. which are due to sampling and computational errors.

-131-
6.11 Summary and Conclusions

The present chapter was devoted to the modeling of the feed-through sampling-head.

The theory and construction of the sampling head were reviewed. The physical dimensions of

the structure discontinuities were used to formulate an equivalent network for the samplinghead input and output lines. The dimensions of the biconical cavity together with TDR

measurements were used to model the cavity part of the sampler. The dc characteristics of

the diode together with TDR measurements vs. sampling diode bias were used to evaluate

various diode model parameters. The strobing pulse was measured, and the data corrected

for both amplitude and risetime.

All of these sampling-head parameters were simulated in the form of a digital computer

program which was run to obtain a theoretical estimate of the sampling-head step response.

The program was implemented in two steps because of the large number of elements and information required to simulate the sampling system.

The step response was found to vary with the diode reverse bias; the higher the reverse

bias the faster was the step response, within limits. The step response at the manufacturer

suggested bias and at the bias for the fastest step response was obtained.

The homomorphic deconvolution transform developed in chapter 2 was slightly modified

in order to be used to deconvolve the sampling-head response contribution from the output

of the sampling oscilloscope. Deconvolution of the observed waveform provides an estimate

of the actual (true) waveform at the input port of the oscilloscope.

-132-
SIGNAL OF

INTEREST

^ OBSERVED

y(t) (RECORDED) * SIGNAL

y(t) = x(t) * h(t)

Y(e^^) = X(e^^) • H(e^^)

Figure 6.1 Oscilloscope Response Effect

on Measured Signals.

50 n

TERMINATION

Figure 6.2. Feed-Through Sampling Oscilloscope

as Used in TDR Measurements.

-133-
Figure 6.3. Two-Diode Sampling Circuit.

>

i

-134-
DISPLAY

TIME

Figure 6.4. Sampling Process.

-135-
-136-
TRAVEL TIME

CONDUCTOR (SAMPLING PULSE)

INCIDENT

STROBE PULSE

(INPUT)

REFLECTED

FORMED STROBE

(SAMPLING PULSE

Figure 6.6. Strobe Pulse Shaping.

-137-
BIASING

VOLTAGE

Figure 6.7. Diode Pack.

-138-
OUTER CONDUCTOR INNER CONDUCTOR

INNER DIAMETERS (rm)

LENGTHS (mm)

43.4416

23.6804

16.2890

8.9586

4.7193

0.0000

4.7193

8.9256

16.3068

23.6652

6.9977

i

= 19.7612

4 9835

7 3914

2 8575

7 3457 , 2 0320

4 2393

2.8727

2 0447

4 2062

2 8448

7 3812

4 9962

7 3584

7.0104

75.5904

ro

CO

99.3064-

OUTER DIAMETERS (mm;

LENGTHS (mm)

3.0429

18.9433

2.1590-1 .9588

2.1636

7 .6149

1 .2413

7.6327

0.8890

4.2850

0.6604

1 .1 557 1 .2954^^>—

J

1.3056 k \ 2.2657-1 .9355 —

0.6604 /

1 .1379

0.8814

4.2977

1 .2446

7.6454

2.1654

7.6200

3.0218

25.5473

2.0244

2.0269

3.0231

47.2211

J

J

43.4416

24.6253

16.8834

• 9.2507

4.9657

3.8100

- 0.0000

3.8100

4.9479

9.2456

15.8910

24.5110

50.0583

51 .0718

52.0852

99.3064

Figure 6.8. Sampling Head Dimensions

-139-
INPUT

00607

00884 —

>

^ .00803

I .01016—

' .00246

^

.00349

.00349 —

,00246 —

,01016—

, 00803 —

18.9433

0.8179

6.7970

0.5944

7.0383

0.2921

3.9929

0.2464

0.2286

3.9776

0.3200

7.0612

0.5842

6.7742

0.8382

25.6489

1.0135

1 .0135

47.2211

2kQ

BLOW-BY

RESISTOR

I .1

Figure 6.9. Tapered Line Equivalent Circu

-140-
a. With Dielectric.

With no Dielectric.

gure 6.10. TDR on Cavity with no Diodes.

Vert. 10 mV/div (p = .04 2 5/div)

Horiz. 100 ps/div

-141-
<!) = — = .0259 VOLTS (AT 300° k)

Figure 6.11. Diode Electrical Model.

mA

Figure 6.12. Circuit Used to Measure Static

Characteristics of a Sampling Diode.

-142-
Figure 6.13. Didoe Static Characteristics.

-143-
-144-
a. Horiz. 1 ns/div, Vert. 5 0 mV/div.

b. Horiz. 10 ps/div, Vert. 50 mV/div.

Figure 6.15. TD Pulse.

-145-
O

C

to

4-)U

•H

-PU3

O

to P

O

uH

Cm

146-
a. Two Diodes

b. One Diode

Figure 6.17. To be continued on next page.

-147-
c . One Diode and an Open Diode

Figure 6.17. TDR on Diodes.

Vert. 5 mV/div

Horiz. 100 ps/div

-148-
Figure 6.18. Sampling Gate Impedance Z /2

.

-149-
-150-
a. V = 0.0 Volt.

b. V = -1.0 Volt.

Figure 6.20. To be continued on next page.

-151-
Figure 6.20. TDR vs. Bias.

Vert. 5 mV/div

Horiz. 100 ps/div

-152-
-153-
CPcH

>

u

o

5-1 O

cu D

4J (U

<D CO CO

rH CU

-p 0) c

0) o

6 J-l H -pCO

<Da Q) x: w -p

CM

CM

0)

•H

<:

cl

CO

-154-
Figure 6.23. Observed Strobe Pulse.

Vert. 1.581 v/div.

Horiz. 20 ps/cm.

-155-
-156-
U

-157-
a.

a

CM 00 vo

LD CM tn o Lf) o cr» CO Ln oLn

C\J OCO

ea- in

VD CM

cn o

^ CO vo

CM — >—

oo o o

o

cri o o <:i- LT) tn

C71 o «:f ID

o cri ^ o o in LD

CMo

LDCMO

cn

o

in

cn

CO

CTi oun

O

COo

O

5 ©0©©©©.®

1

Figure 6.26. Simplified Network for Structure

Step Response.

-158-
> Q\Z

UJ

CO

0-1 roo

COrM

I—I oXQ

coo

CO

LU

CMO

=i H

fa0

CO 0wc0

(0

0)a0

-p

CO

0)}^o

^^

U3u

-15 9-
Figure 6.28. Sampling Network.

-160-
Figure 6.29. Sampling Efficiency and Feedback.

-161-
-162-
-1 6"^-

-165-

-167-
a. Low Reverse Bias (-1.46 Volt).

b. Bias for Optimum Sampling (-1.63 Volt).

Figure 6.36. To be continued on next page.

-168-
c. High Reverse Bias (-1.76 Volt)

.

Figure 6.36. Effect of Diodes' Reverse

Bias on Sampling.

Vert. 50 mV/div.

Horiz. 20 ps/div.

-169-
-170-
o\

cru

IT) 04CDO

CO

rsi I—

a:oXQ

CMo

CO

UJ

4J

Cfl (U pn

(d fa

<u

-POm

Q)

UlCO

cuw

0)

CU • (U H

4-) I

T3

O>

CO •H

CQ

cnC0

CO

00nH

fa

-171-
-Aa\

CO2:

LLi

10

in

O

N*—

«

o

X

o\

2

I

oCO

UJ

o

COc

o

w

CO

CO m

p - g CDo

73

IC

•H

Q

4JH

O

>

(U

5-1 13 •H

-172-
-173-

-175-
-176-
-177-
-178-
7. SUMMARY AND CONCLUSIONS

This report was devoted to the study of the theory and application of the homomorphic

transformation to time domain spectroscopy and scattering problems

.

Chapter 1 was an introductory chapter which reviewed the contents of the work presented

in this report.

Chapter 2 started with a review of the convolution relations in both the time and

frequency domains and the need for deconvolution (separation) techniques. The general

class of the homomorphic systems was defined, and their properties were stated showing how

they can be used for signal separations. In particular, the homomorphic deconvolution

transform was developed. The related problems, such as the nonanalyticity of the complex

logarithmic function and the effect of the linear phase component, were presented and

discussed, and solutions to such problems were given. The computational realization of the

deconvolution transformation was made possible in the form of digital computer subroutines

HDT and HDF. Chapter 3 presented several classes of time domain measurement problems in the field

of linear microwave network analysis. The homomorphic deconvolution transform, that was developed in chapter 2, was used in deconvoluting various convolutions in these problems.

The examples that were considered are

:

(1) pure resistive loading along a transmission line,

(2) commensurate transmission line networks,

(3) TDT type measurement and with two reactive discontinuities, and

(4) conventional TDR measurement to determine a reactive load impedance.

The homomorphic deconvolution method was proven to be very effective in separating convolutions. The method was not limited by the overlap of consequent reflections in the time

domain.

In chapter 4 the theory of the time domain dielectric spectroscopy measurements was presented. Conventionally this method of measurement works only for nonoverlapping reflections; i.e., reflections occurring in separate time domain windows. If the reflections

overlap, only approximate solutions may be applied to the frequency domain form of the

signal. By applying the homomorphic deconvolution transformation to the TDDS waveform, a new method was developed which was not limited by overlapping reflections. The new method

was applied to the well known Debye dielectric model. This method eliminated the over- lapping reflections and determined the dielectric parameters to within 0.3% error of their

model values. The new method is a general one and has great utility for time domain measurements on dielectrics since it does not include any restrictions on the dielectric

material parameters or require the assumption of a prior physical model. Furthermore, the new method is not restricted to time domain spectroscopy of only dielectric materials but

has utility for time domain spectroscopy in general such as for magnetic and other phenomena.

-179-
Chapter 5 began with an introduction about the role that the time domain measurement

techniques can play in solving electromagnetic radiation problems. Then the distributed

nature of the expected cepstrum in such problems was discussed and compared with previously

studied discrete cases. Two examples of applying the homomorphic deconvolution transform

to an antenna system were presented, and the transform proved to be very successful in the

removal of the unwanted reflections and scattering components either overlapping or not

overlapping the main signal containing the desired signal information.

Chapter 6 was devoted to the modeling of the feed-through sampling-head. The theory

and construction of the sampling head were reviewed. The physical dimensions of the structure

discontinuities were used to formulate an equivalent network for the sampling-head input

and output lines. The dimensions of the biconical cavity together with TDR measurements

were used to model the cavity part of the sampler. The dc characteristics of the diode

together with TDR measurements vs. sampling diode bias were used to evaluate various diode

model parameters. The strobing pulse was measured, and the data corrected for both amplitude and risetime. All of these sampling-head parameters were simulated in the form of a digital computer program which was run to obtain a theoretical estimate of the samplinghead step response. The program was implemented in two steps because of the large number

of elements and information required to simulate the sampling system. The step response

was found to vary with the diode reverse bias; the higher the reverse bias the faster was

the step response, within limits. The step response at the manufacturer's suggested bias

and at the bias for the fastest step response was obtained. The homomorphic deconvolution

transform developed in chapter 2 was slightly modified in order to be used to deconvolve

the sampling-head response contribution from the output of the sampling oscilloscope.

Deconvolution of the observed waveform provides an estimate of the actual (true) waveform

at the input port of the oscilloscope.

-180|

I

)
8 . ACKNOWLEDGMENT

The first author acknowledges the National Bureau of Standards for providing him with

a Guest worker appointment to work in the Boulder Laboratories under the NBS/University of

Colorado Joint Graduate Program in Electromagnetics Metrology, and the cooperation between

the Graduate Schools of the University of Toledo and the University of Colorado which

facilitated the continuance of the author's dissertation research in Boulder, Colorado.

Also, the first author \rf.shes to acknowledge the invaluable assistance and guidance

of his advisor. Dr. N. S. Nahman (Chief, Time Domain Metrology Program, Electromagnetics

Division, National Bureau of Standards; Professor, Electrical Engineering Department,

University of Colorado; and formerly Professor and Chairman, Electrical Engineering

Department, University of Toledo), who suggested the subject of this report and contributed

in many ways to the completion of this work. As an NBS Guest Worker, the first author had the privilege of doing his research in

close association with J. R. Andrews, Ph. D. , W. L. Gans, M. S., and A. R. Ondrejka, M.S.,

each who contributed a stimulating environment for the author's research. Finally, sincere

thanks go to Mrs. Sandra McCarthy, Miss Sharon Foote, and Mrs. Sheila Aaker who significantly

contributed to the editing and preparation of this report.

-181-
APPENDIX A

THE z-TRANSFORM AND THE DISCRETE FOURIER TRANSFORM

The z-transform is a generalization of the Fourier transform in the case of discrete

time signals (sequences). The z-transform of the sequence x(n), where n is the sequence

variable, is defined as

z[x(n)] = X(z) =

I x(n) z ^

(A.l) n=- <»

where z[ ] is the z-transform operator and z is a complex variable.

The z-transform X(z) does not converge for all sequences x(n) or for all values of z. The region of convergence, i.e., the set of zvalues for which X(z) of a given x(n) con- verges, is very important. It must be defined in association with the transform X(z) in

order for the inverse transform to be uniquely defined. Generally speaking, the z-transform,

eq. (A.l), converges in an annular region of the z-plane given by

0 <_ R_ <

I z I < R^ £ ~ . (A. 2)

For a stable time domain sequence, the convergence region of the z-transform includes the

unit circle |z| = 1. The inverse z-transform is given by

x(n) = 2^ ^ X(z) z""-"- dz (A. 3)

where C is a counterclockwise closed contour in the region of convergence of X(z) encircling

the origin of the z-plane.

If the complex variable z in eq. (A.l) is expressed in the polar form

z = re-J (A. 4)

the z-transform will take the form

00 x(re-^ ) =

I x(n) re-' (A. 5)

n=-oo which can be viewed as the Fourier transform of x(n) multiplied by an exponential sequence

r If r = 1 (|z| = 1) the z-transform becomes identical to the Fourier transform of the

sequence,

X(ej'^) =

I x(n) e-j^". (A.6)

n=-«° For finite duration sequences, with total number of samples N, the z-transform, eq. (A.l), will take the form

N-1 _ - X(z) =

I x(n) z (A.7)

n=0 "1^^-
or

N-1

X(e"''^) = 1 x(n) e

-'"^

, on the unit circle. (A. 8)

n=0

For computational reasons, only a finite number of samples of the z-transforms can be

calculated. Sampling the z-transform at N-equally spaced samples on the unit circle

|z| =1 results in the discrete Fourier transform DFT,

X(k) = XCe^"") , = 2-k/N

, , -i(2TTk/N)n = V x(n)e (A. 9)

n=0 wnere k is a sequence variable that takes on values from 0 to N-1. By this sampling process

the finite duration sequence x(n) is forced into being periodic with period N. Tne transform X(k) will also be periodic with the same period. The discrete inverse Fourier transform DIFT is given by

x(n) =^ Y X(k) e^C2.k/N)n_

k=0

Both tne z-transform and the DFT have the following listed properties even though

these properties will be presented only in the z-transform form. It is to be remembered

tnat tne DFT corresponding relations can be obtained by replacing the complex variable z by

^j(2T:k/N)n operator z[ ] by DFT[ ]. The properties are:

1. Linearity

z[a x(n) + b y(nj] = a X(z) + b Y(z) . (A. 11)

2. Sequence shift

-n z[x(n-n^)] = z ° X(z). ^ (A. 12)

3. Convolution

z[x^(n) * x^Cn)] = z[x^(n)] • z[x2(n)]. (A. 13)

Finally, one additional but very useful property of the z-transform will now be discussed;

it is the multiplication by an exponential sequence. This property is

z[a^x(n)] = X(z/a). (A. 14)

The importance of this property lies in its ability to scale the convergence region by

tne factor 1/a, tnus enabling the inclusion of the unit circle in the convergence region

and, consequently, stabilizing unstable sequences. This method can also be used to remove the transform zeros that may lie on the unit circle giving rise to singularities

upon taking the complex logarithm. This method of scaling the z-transform is often

called the exponential weighting.

-183-
APPENDIX B

COMPUTER SUBROUTINES HDT AND HDF

SUBROUTINE HDT (X.H.N. NSH)

Ctttt T^C HOMOMORPHIC DECOHUOLUTION TRANSFORM

C**** FOR TIME DOM SIGNALS

CmPLEX X<1).H<1).CLX,CC

C  TAKE T^E FOURIER TRANSFORM

CALL FFT (X.N. .TRUE. > C** ** CALL SUBROUTINE HDF

CALL HOF (X.H.N.NSH)

C**.t* INl€RSE FOURIER TRANSFORMS OF SEPARATED COMPONENTS

CALL FFT (X.N. .FALSE.

)

C^ FFT (H.N. .F^^SE. ) RETURN

END

SUBROUTINE HDF (X/H.N.NSH)

Ctttt Jm HCM3M0RPHIC DECONUOLUTION TRANSFORM

C**** FOR FREQ DOM SIGNALS

COMPLEX X< 1 >. H( 1 >. CLX. CC

PI=4.^TAN<1.0)

N2=N/2-t-l . . C**** CONTINUOUS COMPLEX LOGARITHM

C=0.0

ARX=0.0

00 2© L=1.N2

ARS=ARX - CC=X(L)

cm, CLG (CC.CLX)

ARX=AIM^^CLX>

AR«*m-^^

IF <m.GJ.'^iy c=c-2.«Pi

IF (AR.LT.-PI) C=C+2.«PI

CC3OIPLX<0.0.C)

X<L>=CLX+CC

20 CONTIM^

Ctttt LINEAR PHASE ELIMINATION

NSH=AIM^^X(N2))/PI

TYPE " SEQ SHIFTED". NSH." SAMPLES"

00 45 L=1.N2

45 X(L)»X<L>-CMPLX<0.0.2.*<L-n«NSH*PI/H)

-184-
K3=N2-2

DC 25 L=1>K3

25 X<N2+LM:0NX<X<N2-L>) C*m INUERSE FOURIER TRANSFCfRM

CALL FFT <X/N. .FALSE. > TYPE " CEPSTRUfI"

Ctttt SEPARATION AhiO IKUERSE PROCESSING

CALL SEP CX>H/N.NSH)

RETURN

END

SUBROUTIKE FFT (X.N.FI)

C**** FAST FOURIER TRANSFORM ^ FI=.TRUE.

C  INUERSE TRANSFORM : FI=.FALSE.

CQ^SPLEX XCN?.U>W>T

LOGICAL FI

EH=N

M=#:^CG< EN >/?LOG< 2 . >+0 .

5

N2=N/2

N1=N-1

J=l

DO 7 1=1. Nl

IF (I. (2. J) GO TO 5

T=X< J

)

>« J >=X< I ) X< I )=T

5 K=N2

6 IF <K.GE.J) GO TO 7

J=J-K

K=K/2

GO TO 6

7 J=J+K

PI=4.tATAN<l .0)

DO 20 L=1.M

L2=2** L

Ll=L2/2

IF <Ll.EQ.i) GO TO 8

U=<i..0.)

D=L1

IF < .NOT.Fn D»-0

W=CriPLX( COS<PI/D).-SIN<PI/0>>

8 DO 20 Jal.Ll

DO le I«J.N;L2

IP1«I-H-1

T»X<IP1

)

IF <J.^€.1> T»T*U

X< IPl >=X< I )-T

10 X< I )=X< I HT

IF (Ll.NE.n U«U*W

20 CONTINUE

IF <FI) GO TO 40

DO 38 I«1.N

30 XCI>=X<I)/EN

40 RETURN

END

-185-
SUBROUTINE SEP <X.H.N.NSH)

Ctttt SEP^JRATION OF CEPSTRUM COMPONENTS AND INUERSE PROCESSING

COMPLEX X<n.H<l).CC

DI^ENSION NS<5).NF<5>.RX<1024)

LOGICAL HPF.IS

00 48 L-l.N

40 RX<L>=REAL<X<L>>

Ctttt LINEAR FILTER < SEPARATION PROCESS )

30 TYPE " READY FOR SEPARATION " ACCEPT "NUf^ER OF ELIMINATED REGIONS «? 'SN2R

DO 5 1=1. N2R

ACCEPT "ELIMINATE POINTS STARTING AT •SNS<I)

5 <^CEPT " AND Ef^ING AT %NF<I>

NS<NZR+1>=N+1

HPF=.TRUE.

NS1«NS< 1 )-l

90 DO 10 L~1.NS1

X<L>Cff=>L>«R>«L).0.0>

10 IF <HPF) X<L)«<0.0>0.0)

DO 16 1=1. NZR

NSIaNS< I

)

NFIaNF<I)

DO 12 LsHSI.NFI

X<L>=8.0

12 IF X<L)«RX<L>-X<L) NFIls^dHl

NSI1«NS< I+l >-l

DO 14 LaNFIl/NSn

X<L>«aiPLX<RX<L).0.0)

14 IF <hPf> X<L)«»<0. 0.0.0)

15 CONTINl€ OPm IWERSE PROCESSING ' FOURIER TRANSFORM

CMl. FFT <X.N. .TRIE. > Ctttt RETAIN LINEAR PHASE AND COMPLEX EXPONENTIAL

00 20 L«1>N

CC=CfW« 0 . 0. 2 . «< L-1 )*NSH*PI/N )

IF ( NOT.^PF) X<L)«X<L)+CC

X<L>«CEXP<X<L))

20 COHTIKE

IF < NOT.HPF) GO TO 99 Om* HPF F^Q DOM SIGNAL

DO 70 L^l.N

70 H< L>=X<H > Ontt REPEAT FOR LPF COMPONENT

HPF=.Fj^SE.

GO TO 9®

99 ACCEPT "sWTHER SEP TRIAL ? -laYES. 0-NO MS

IF <IS) GO TO 30

RETURN

END

-186-
APPENDIX C

COMPUTER PROGRAMS TDNA AND SMPL

C.l Program TDNA

C**:*:* PROGRAM TDNA ' TI^E OOM NETWORK ANALYSIS

C**** THIS PROGRAM CAN HAmLE ANY OF THE FOLLOWING NETWORK ELEMENTS • Ctttt 1= CUR SRCE.. R= RES. 0= DIODE <NLR). C^ CAP. N' NL CAP.

C**** V- ULTG SRCE. L= M. T= TRANSM LINE

DI^ENSION CK651. n >.C1T< 11 >.C2<651. 11 ).C2T< 11 >.R0< 11 ).NT< in

DIMENSION IBNCH< 7. 100 >. ELUALC 100 ). NOOOUTC 2. 30 ). UOUT< 30

)

DIMENSION TIME<651>.TDU<651.1).TDUN<1)

DIMENSION G<35.35).C<35).U<35>.USAU<35).NU<10>

DIMENSION UPLOK 651 . 26 > COWON mMPTS.TMIN.TMAX

NDIM=35

M=0

C^^M^^ READ If"^UT DATA

OUJ- RB^IBNCH.ELUAL.^CL.NN.NODOUT.NOUT.R0.LL.NU)

T=TI*ttX<'i'^Jf^TS

NU1=NU<1) $ NOUTNU«NOUT-»+IUl $ NUPl-NUl+l

LLL=2^ $ NOUTLL«NOUT+LLL

NUMPTS#^UMPTS-H

DO 2 Jal.NUMPTS

TDU<J.1)=0.2S

2 TIME<J)=<J-1)*T

TDU<1.1)=0.0

IS=1

CALL PLOT (TIME.TOtXl.D.NUMPTS.ieWINPUT US T)

Ctttt ADJUST ELEMENT LIST

CALL PREPAR<IBNCH.£LUAL.NEL/m.T.NT.R0>

DO 18 J«1.NN

USAU<J)a0.0

18 U<J)»0.0

DO 2© L«1.LL

C1T<L>«0.0

20 C2T<L>e.0

WRITE <6.89) NEL/W.U-.NWTS.Nyi.NCSJT/NOUTW/TOfrLL.NUPl/LLL

l«ITE <6.92> <<<imCH<JJ.n>.JJ»1.7>.ELUAL<n>)/n-l/Na.>

DO 50 N«i.NUMPTS cm* SET UP NODE EQUATIONS

DO 22 n«i.is

22 TDUH<n>=TOU<N. ID

24 CALL NOOEQ < IBNCH.ELUAL.NEL.G.C.U.NN.NDIM.USAU.T1ME<N)/T.TDUN/

X C1T/C2T)

IF <N.GT.1.0R.M.GT.l) GO TO 28

tRITE <6.92> <<<IBNCH<JJ.II).JJ«1.7>,ELUAL<II)>.II-1.NEL)

DO 26 11=1. NN

26 WRITE <6.91> <G< II. JJ). JsJ=l.NN).C< II ).U< II )

28 CONTINUE

C«m SOLUE NODE EQUATIONS

CALL SLNEQ <NN.NDIM.G.C)

-187-
Ctt^t y<J)-C<J> IS THE INCREMENT IN THE JTH NODE UOLTAGE

ERR=0.®

DO 30 J=1.NN

ERR=€RR-K U< J >-C< J > )*3|c2

30 U<J)=€<J>

Cmt TEST COHUERGENCE CRITERION

IF<ERR.GT.l.E-6.AND.ri.LT.15)G0 TO 24

M?ITE <6.93) N.U<8>

CALL OUTPUT<NODOUT,NOUTLL/U.UOUT>

00 32 IIsl/NOUT

32 UPLOT<N.II)«U0UT<n)

DO 34 JJ«!/NU1

n»wuT*jj $ NUjaNU<jj+n

34 UPLOTCN/IIMKNUJ)

TO 36 Lal/LLL

II»NOUTNV+L $ JJ-NOUT-H.

36 UPLOT<N.II>»UOUT<JJ) Cmt STORE TRANSMISSION LINE CURRENTS

00 38 L«1/LL

IL«N0UTNU+2*L-I

C1<N.L><^PL0T<N. IL>^^L>-C2T<L>

38 C2<N.L>«UPL0T<N. IL+1 VR0<L)-C1T<L)

Ctttt STORE NODE UOLTAGES FOR UPDATING CAPACITORS

DO 40 Jsl.NN

40 USAU<J)=U<J>

M=0

DO 45 L=1.LL

IF <N.LT.NT<L>) GO TO 50

Ontt UPDATE TL DELAYED CURRENT SOURCES

^«^i=N~HT<LHl S IL«N0UTW'»-2$L'-1

C1T< L )=UPLOT< NNTl . IL )/R0< L )+Cl< NNTl . L

)

45 C2T<L>=W-0T< NNTl. IL+1)/1^LHC2< NNTl /L)

50 C43NTINIE

00 55 II=1.N0UTNU

ITITL=IT<n.NOUT>

55 m± PLOT <TIfCUPL0T<l.II),Mf1PTS.ITITL)

I^ITE <6.94> TMAX

IF <NOUT.EQ.0> GO TO 65

WRITE <6.95) <JJ.<N0D0UT<II/JJ)/n«1.2>/JJ«l.N0UT>

DO 60 Nal.NUMPTS

TNR=TIPE<NVTMAX

68 WITH <6.91) TNR.<UPLOT<N/II>.n«l.NOUT>

65 IF <NV!.EQ.0) GO TO 80

IF <NOUT.hE.0> WRITE <6.94> Jfm

WRITE (6/96) <NU< ID. n«2.NUPn

DO 70 N-l.NUMPTS

TNRaTIMECNVTMAX

70 WRITE <6.91) Th«.<UPLOT<N.n-^HaiT).II"l/NUl)

80 WRITE <7.97) <UPL0T<N.N0UT+1 >.N»iiNU«PTS)

89 FORMAT (1016)

90 FORMAT (8F10.0>

91 FORMAT <1X/10F13.7)

92 FORMAT < IX. A4/6I3/E15.5>

93 FORMAT <I7.E1S.9)

94 FORMAT <1H1.« Tm< » «/E12.5/>

95 FORMAT <8X.«T/0rMAX «.8<« BW/ 12.» 0ET«.2I2>>

96 F^WAT <^ T/TMAX *.9<* N0DE»/I3))

97 FORMAT <5E16.9>

STOP

END
c.2 Program SMPL

Ctttt PROGRAM SMPL = SAMPLING-fCAO STEP RESPONSE

DIMENSION IBNCH<7.42).ELUAL<42)

DIMENSION G< 16. 17 ). C< 16 U< 16 >. USW< IS > DIMENSION TDU<500.2>.TDUN<3).TOC»S<500>

Dlf^SION CK 500 >.C2< 500 )..UPLOT< 500.2)

M=0

Cttt^ READ INPUT DATA

TYPE "INPUT NEL.NN..NPTS,NUf^TS.NT/T.R0.NBB.Nll.N2i>N12.N22"

ACCEPT N£L.NN.NPTS.NUMPTS.NT>T.R0.N^,Nll.N21.Ni2.N22

ACCEPT "INPUT = NBC^..NC..U0.T0UN<3) ".NBGN.NC.U0.TDUN<3)

READ (13.92) << IBNCH< II. JJ>.. II=1.7).ELUAL': JJ). JJ-l.NEL)

WRITE <10.92) <<IBNCH<II.JJ).II=1.7).ELU^<JJ).JJ=1.NEL)

READ < 13.97) <TDU< J..2). J=1.NPTS)

READ < 13.97) <TIXK J. 1 ). J=1.NUMPTS)

TIXi2F=TDU<NPTS.2)

DO 4 J=i.NUMPTS

TDU< J. 1 )=TDU< J. 1 )/25 .

0

IF <J.GT.I^TS) TDU<J.2>«TDU2F

TDU< J.2)=TDU< J.2)/R@

4 TIXJS< J)=TDU< J.2)

D1IU=£LUAL<NBB)

D2IU=ELUAL<NB8+1)

ACCEPT "FEED SACK FACTOR = ".F^

NSTP=i

NPTS=250

DO 80 NSPE=NBGH.Wjr^TS.NSTP

W3«NSPE-I^TS

IF <ND.£Q.0) GO TO 16

8 J«1.ND

8 TDV<J.2)«TDUS<1)

DO 10 Jal.NPTS

10 TDU<J+ND>2>«TDUS<J>

16 CONTINUE

DO 18 J«1.NN

USAU< J>=0.0

18 U<J)=6.9

ELU!^< )Oi lU-^TD*^ 3 )*1 . E5

ELUAL< hm-i-i >=02IU-TDW< 3 )*1 . ES

ciT»0.a

C2T«0.0

DO 50 N=i.NSPE

C**** SET UP NODE EQUATIONS

TDUN<l>»TDU<N.l)

TIX^2)=TDU<N.2)

TIME=<N-i)*T

24 CALL NOKQ < IBNCH.ELUAL.NEL.G.C.U.NN.NDIM.USA<J.T2f€<N)>T.TDUN.

X C1T.C2T)

CW** SOLUE NCOE EQUATIONS

CALL SLHEQ <NN.NOIM.G.C> cm* U<J>-C<J) IS THE INCREMENT IN THE JTH NODE UOLT<^

ERO=0.0

DO 30 ^l.m

ERO=€RO+< U< J >-C< J ) >**2

30 IXJ>=C<J>

f1=fHl

-189-
amt TEST CONUERGENCE CRITERION

IF<ER0.GT.1.E-6.AND.M.LT.15)G0 TO 24

W.OT< N. 1 )=U< Nl i )-U< N21 > UPLOK N/ 2 >=U< N12>-U< N22 )

CUtt STORE TRANSMISSIC»^ LI^€ CURRENTS

Ci< N )sUPLOT< N. 1 )/R8-C2T

C2( N >=UPLOT< H. 2 )/R0-ClT

CUtt STORE NODE UOLTAGES FOR UPO#^TING CAPACITORS

DO 48 J^i/NN

40 USAU<J)sU<j)

IF <N.GT.l) GO TO 42

UCiIaU<2)HKl)

UC2I»U<a>-U<9)

42 Ma®

IF (N.LT.NT) GO TO 50

Ctttt UPDATE TL DELAYED CURRENT SOURCES

NNTl^N'-NT'^l

C1T=UPL0T<NNT1. 1 )/R0+Cl<NNTl

>

C2T=UPL0T< NNT 1 . 2 )/TOK:2< NNT 1 > 50 CONTII'^^

USC=< U< 8 MX 9 ) )-UC2I-< U< 1 )-U< 2 ) )-UCl I NC=hC+l

TOUN< 3 )=TDUN< 3 )+FBF:*sUSC

NCT=< NC-1 >*T/1 . E-12+ .

5

U0=U0+USC

mUE <10..93) NC.NCT.U0/TaiN<3)

80 CONTINUE

92 FC^MAT <1X.A4.I6.5I3.E15.5)

93 FORflAT <1X.I5.5X.I5.2X>"PS".2E17.6)

97 FORMAT <5E16.9)

STOP

Eh©

-190-
C.3 Subroutines Used in TDXA and S>fPL

SLBROUTINE RE^^< I^O^. ELUAL. ^CL. NN. NODOUT. HOUT. R^. LL. ) DIMENSION IBNCH;?. 1 ^^ELUALC 1 >.NOD0iIT<2> 1 )>DUmV< 15).Re< 1 > INTEGER TA.NUC 1 ). IU25).NUT( 10);NUA<9)

COMMON NUt1PTS;TFMIH.TFmX

DATA IBLAN*</1H ^ ^ NUi^^/lHA. IHe. IHC. IKO. 1H£. IHF. IHG/ 1HH> IHI/

C**** REM) AND WRITE NAUE CARD

READ <5..50) (OmMYC J). J=i. 15?

l^ITE (6.52) <DUmY( J). J=l. 15)

L=8 $ NN=e * N£L=5 $ NCHJT=0

Ctttt R£f€ hCXT ELEMENT

READ (5.54) < IBNCH< J. NEL). J=l . 5 ). ISGN; lOUT. IS. < DLWYC J ). J=l . 2).

X <N<vT( J). J=l. 10)

C**** READ FIRST CHARACTER CF NAf^ <=ELEr£NT TYPE)

BACKSPACE 5

READ <5.5€) TA

OPm DECODE ELEflENT TYPE

NTYPE=NOX€< TA ) IF <NTYP£-9) 20.35. 10

Ctttt SAC€ GEN^ERAL INFORMATION CARD DATA

10 N!J1PTS=IBf^H<4.N£L)

TFMAX=DLfim'< 1 ) $ TFNIN«CX«MY<2)

IBNCH< l.h€L)=4H

NEL=NEL-1

DO 15 J=1.10

15 NU<J)=NJT<J)

NU1=NU<1)

IF <NUl.e3.0) GO TO 5

fcRITE <6.58) <NivKJ+l).J-l.NUl)

WRITE (€.60) J).J«1.NU1)

GO TO 5

20 ELUALCNELXBJ««Y<1) Opm STORE TYPE OF ELE^E^f^

IBNCH<6.Ha-)«NTYPE

IBNCH<7.S£L)==0,0

IF (CNTYPE.EQ. 1 ,0R.NTYPE.EQ.6).AN0. IS.NE.0> IBNCH< hSL >«IS*4 cm* STORE OUTPUT NODE NUMSERS

I0UT=NOXE< I OUT)

IF <I0UT.^€.6) GO TO 30

NOUT=NOUT+1 cm* CHECK SIGH OF CBJTPUT UOLTAGS

IF <I9>I.E]3.IBLANK) GO TO 25

NOOOUT< 1 . NOUT >= iaCH< 5 > hCL)

NOCXXIT< 2. NOUT >=IBNCH< 4. NED

GO TO 30

25 HOOOUTC l.N0UT>*I^CH<4.hCL)

NOOOUTC 2. NOUT )« I BNCH< 5 . NEL

)

cm* STORE HIGHEST NODE NUMBER

30 NN=*W<9<NN.IBNCH<4.fCL).IBNCH(5.h€L)) cm* READ OUTPUT NODES OF TRANStllSSION LINE

IF <IBNCH<6.NEL).NE.8) GO TO 5

L=L+1

READ <5.62> N3.N4.R8<L)

hfW1AX0<hM.N3.N4)

-191-
IBNCH<2.NEL)=N3 $ IBNCH<3.he.>«N4

GO TO 5

35 IBNCH<l/hEL)«4H

N£L=NEL-i

LL=L

DO 40 Lsl^LL

liL=ll<L) $ IL»NOUT+2:5cL-i

N00ajKl/IL>«IBNCH<4>nL> $ NOOOUT< 1. IL-H )«IBNCH<2. IIL)

N000UT<2.2L)«IBNCH<5.nL> $ N0D0UT<2> 2L+1 >«IBNCH<3. UL>

40 CONTINUE

URITE <€/64)

L«0

QQ 4^ Jsa I , NEL

URITE <6>€6> ieNCH<i/J>.IBNCH(4/J>/IBNCH<5.J>^ELMAL<J>/

X 2BNCH<3.J)/IBNCH<2/J>

IF <IBNCH<6.J).EQ.8> WRITE <6.68) IBNCH<2. J). IBNCH<3. J).R0<L-#-l )

45 CONTINUE

RETURN

50 FC^^IAT < 15^)

52 FORMAT < 10< IH*). iS^H. !0< 1H«))

54 FORMAT <A4. A1.A4. 14. 13. 1X/A1.A1> I1.2E15.5/ 10I3>

56 FORMAT <A1

>

58 FORMAT OUTPUT NODES AND THEIR CORRESPONDING ALPHABETIC

X *NA^€S USED IN MICROFILM PLOTS*/. 1X.9<2X. I3>>

60 FORMAT <iX.9<4X.Al))

62 FCm^T <10X.2I3.3X.E15.5>

64 FCFMAT <//* BRANCH h03E ELEMENT BRANCH CON-*/

X * NAME NOS. UALUE NAME TROL«)

66 FORMAT < IX. A4.2I3.E15.5. IX. A4.3X. Al

>

68 FORMAT <5X.2I3.E15.5)

END

SUBROUTIfC PREPAR<IBNCH.ELUAL.NEL.NN.T.NT.R0>

DIMENSION IBNCH<7.1>.ELUAL<1).NT<1).R0<1>

COMMON K01PTS.TMIN.TMAX

DATA IBLANK/IH /. IREF/lHI/ C*m mTCH CCOT50LLED Pm CONTROLLING ELEMENTS

00 2 I«|.NEL

IF <IBNCH<6.I).NE.1> GO TO 2

IF <IBNCH<2.I>.EQ.IBLANK> GO TO 2

IF <IBNCH<2.I).EQ.IREF) GO TO 2

DO 1 J^l.NEL

IF <IBNCH<3.I>.NE.IBNCH<1.J)) GO TO 1 cm* STORE CONTROLLING NODE NUMBERS FOR USE IN NOOEQ

IBNCH< 2. I )=IBNCH< 4. J

)

IBNCH< 3. I >«IBNCH< 5. J

>

GO TO 2

1 CONTINUE

2 CONTINUE

f€Ll=NEL

C**«* MODEL INDEPENDENT UOLTAGE SOURCE BY

C GVRATOR AND INDEPE^«DENT CURRENT SOURCE

00 3 I»l/NELi

IF <I8NCH<6.I).NE.S> GO TO 3

-192-
IF <iaCH<2.I).NE.IBUW) GO TO 3

GYRATR<I8HCH.ELU^.N£L>NN.M.0.1.8) cm* MO»v€ CHfii^ TYPE OF IbCEFEHDEHT SOJRCE

IBNCH<4.I)=NN

3 CONTINiJ€ Om* f-QOEL INDUCTOR BY GYRATOR

^CAPACITOR

DO 5 I-l.NELl

IF <ISHCH<S. n,NE.7> GO TO 5

C^^ GYRATR<ISNCH.ELUAL.hCL.NN/M.@;l.e) cm* flOUS A^C REPLACE INDUCTOR WITH CAPACITOR

IBNCH<4.I>«NN

IBNCH<5.I)»e

IBNCH<6. I )=4

5 CONTINUE

C*»* MODEL SHORT CIRCUIT BY A GYRATOR Afd OPEN CIRCUIT

DO 6 >l.hCLl

IF <IS^>K6.I).NE.2) GO TO 6

IF <EL<^CI).NE.0.0) GO TO 6

SCCT <IBNCH.ELUAL.h€L.h*^;I)

6 COfTINUE

C*«* REPLACE TRANSMISSION LINE BY ITS MODEL

L=8

1X3 7 I=l.hELl

IF <I&CH<6.I).h€.8) GO TO ?

L=L-»-l

NT<L)=£LUAL<I VT+e,5

CALL TRLIN < IBNCH. ELUAL. NEL> I . Re< L )

)

7 CONTINLE

Ctttt MODEL NONLINEAR CAPACITOR BY ITS COMPANION MODEL

DO 8 I«i.NELl

IF <IBNCH<6.I),hE.5) GO TO 8

Tl=l./T

CALL GYRATRC IBNCH. ELUAL. NEL. NN/ I . Tl . 1 . 0

)

IBNCH<7.hEL-l >=1

cm* CHAM3E TO UARIABLE CIRRENT SOURCE

C NOTICE SIGN CHANGE IN hODEQ

IBNCH<3.I)«NW

IBNCH<6.I>«1

I»CH<7.I)»1

C**** ADD UARIABLE COhOUCTANCE AND CURRENT SOURCES

NEL«NEL+1

IBNCH< 1 . NEL >«IBNCH< 1 . I

>

IBNCH<2/h€L>«lH

IBNCH<3.NEL>-I

IBNCH<4.h£L>«NN

IBNCH<5/NEL>»e

I8NCH<6.hCL>«2

IBNCH<7.^€L)=1

NEL-h€L+l

IBNCH< 1 . NEL >«IBNCH< 1 /

n

IBNCH<2.h€L>»lH

IBNCH<3.NEL>»NN

IBNCH<4.NEL>=NN

IBNCH<5.NEL>=0

IBNCH<6.NEL>=1

IBNCH<7.NEL>=2

8 CONTINUE

RETURN
SUBROUTIfC NOOEQ<IBNCH.ELUAL.^CL.G.C.U.^#^>NOIM.USW.TI^fE.T.TDUAL.

X CiT.C2T)

REAL LAMDA

DI^E^^SION IBNCHK 7. l >,ELUAL< 1 ).TOUAL< 1 ).C1T< 1 ).C2T< 1

)

DIMENSION G<N0II1. 1 ),C< 1 ).U< 1 ).USAU< 1)

COmON NUMPTS..TI1IH.Tf1AX

DATA IK-ANK/IH INIT/9// IREF/lHI/

DATA CSAT/'2.76E-ll/'.LAMDA/'38.6i/'

C^^c*** ZERO WOE mTRIX Am CURRENT UECTOR

DO 5 K=l.m

C<K)=0.0

DO 5 j=i.m' • 5 G<J..K>=5@.8

L=0

00 80 J==1.NEL

Om:t Nl AM) N2 ARE NODE WJ-SERS

N1=IBNCH<4.J>

N2=IBNCH<5.J)

BU=e.0 $ BUN=0.0

IF <N1.EQ.0) GO TO 10

B«v>=U<Ni> $ BUN=USAU<N1)

10 IF <N2.£Q.0) GO TO 15

BU=BU-U<N2> $ BUNaBW-USAU<N2)

15 CONTINUE

CtUt DETERMINE ELEHENT TYPE

NTYPE=IBNCH<6.J>

GO TO <2®.25/70.7S>.NTYPE Ctm TEST TO SEE IF INDEPENDENT CURRENT SOURCE

20 IF <I»CH<2.J).EQ.IBUW) GO TO 35

C*t«« MODIFY NONLINEAR CAPACITOR MODEL AT DC

IF <TIME.EQ.0.0.AND.IBNCH<7.J).EQ.l) GO TO 8® Omt LOAD UOLTAGE CONTROLLED CURRENT SOURCE INTO G

CfiLL UCCS<Nl.N2.IBNCH<2^J)/IBNCH(3>J>iELUAL<J>^G>N0IM>

GO TO S© Omt LtM> CQf«XTANCE INTO G

25 IF <IBNCH<7.J).EQ.0> GO TO 30

ICNL«IBNCH<3>J>

NC1=«IBNCH<4.ICNL> $ NC2«IBNCH<3. ICNL>

8MC=0.0

IF <NC1.ME.0> BUC«U<NC1>

IF <NC2.f^.0> BUC=BUC-U<NC2>

ELUAL<J>=CNL<BUC)

30 COM5=i.0/€LUAL<J)

CALL BMCH<N1.N2.C0ND.G>NDIM)

GO TO 80

35 IF <IBHCH<7.J).GT.4> GO TO 60

ICNT=ie^CH<3.J>

GO TO < 65. 40. 45 . 50. 55 IBNCfK 7/ J HI cm* MCS3IFY NLC MODEL AT DC

40 IF <TIME.EQ.0.0) GO TO 8® C*m CmHGE. SIGN AS MENTIONED IN PREPAR

ELUAL<J>«-VSAU<ICNT)/T $ GO TO 63

45 EL«M.<J>aeUC-CONO«U<ICNT) $ GO TO 65

58 L=L*1 $ ELUAL<J)«C2T<L) $ GO TO 65

55 ELUAL<J>sCiT<L> $ GO TO 65

60 IS«IBNCH<7.J>-4 $ ELUAL< J)»TDUAL< IS)

65 CALL SRC£<N1.N2/ELUAL<J).C)

GO TO^

Ct$$t DEFINE DICDE PARAMETERS cm* ^m.Y INITIAL DIODE IF ON FIRST ITERATION

70 IF <INIT.EQ.0> BU»€LUAL<J)

-194-
C3*» IMPOSE HEURISTIC LIMITATION OF .8 UOLTS ON DIODE BU

BU=A«IN1<BU.0.8)

DEX»0.0 S BUJl}«BU*LArClA

IF <SULMD.GT.-6?S,@) C€X»EXP< BULMD

)

GO«CS^T«IJVtM«DEX

CUR-CSAT*< OEX-1 . @ >-iU*0 cm* LOAD (2) AND CUR INTO G C

C<^ BHC>4<N1.N2/GD/G.N0IM)

CALL ^CE<N1.N2.CUR.C)

GO TO 88

Ctttt FORM CJ^AHIOH flOOEL OF CAPACITOR; SKIP IF DC

75 IF <TI?«.EQ.8,8) GO TO 88

GC«ELU^^< J )/T

CUR=8UN^

CALL ^40KNl.N2.GC.G.NOIf1)

CALL SRC£<N2.N1.CLR.C)

80 CONTIhJLS

INIT=1

f^TURN

END

aJ8R(XJTI^€ S_NEQ<N>M3IM.A.B)

Ctttt SOLUTION OF H LIh€AR EQUATICNS A< I . J )*B< J >«B< I

>

DIMENSION A<NDIM.1)>B<1)

N1=N+1

DO 18 1=1.

N

10 A<I.N1>=B<I>

DO 40 K=1.N

K1=K+1

G=8.8

DO 20 I=K.N

P=ABS<A<I>K))

IF (P.LE.G) GO TO 20

G=P $ L=I

20 CONTINL€

DO 25 J=K. Nl

G=A<L>J) $ A<L. J>«A<f<. J) $ A<K.J«

25 CONTIWJE

P=A<K.K)

DO 38 J=K1.N1

38 A<K. J)=A<K. J VP

DO 48 I»1.N

IF CI.EQ.K) GO TO 48

P=-A<I.K)

DO 35 JaKl.Nl

35 A<I.J)«A<I.JHP*A<K.J)

48 CONTINUE

DO 45 >1/N

45 B<J>«A<J.N1)

I^TLRN

END

-195-
SUBROUTIhE GYRATR< IBNCH.ELUAL.NEL.NN, I.G1/G2>

DIMENSION ELUAL< 1 >/ IBNCH< 7.1)

Ctttt THIS SUBROUTINE GENERATES TWO UCCS'S TO SI^aJLATE GYRATOR

N1=IBNCH<4.I) * N2*IBNCH<5/n

NN=NN+i

Ctttt ADD FIRST UCCS

IBCH< 1 . ^€L >«IB^CH< 1 .. I > IBNCH<2.hEL>=NN

IBNCH<3.hEL>=0

IBNCH<4.NEL)=N1

IB^CH<5.hEL)=N2

IBNCH<6.NEL)=1

IBNCH<?.NEL>=€

ELUAL<NEL>=G1

Ctttt ADD SECOND UCCS

^CL=NEL+l

iaCH( 1 > f€L )=IBNCH< M> IBNCH<2.NEL)=N1 . IBNCH<3/NEL)=N2

IBNCH<4.NEL)=0

IBNCH<5.NEL)=NN

IBNCH<€.NEL>=1

IBHCH<?.NEL)«0

ELUAL<NEL)=G2

RETURN

END

SUBROUTINE SCCT< IBNCH.ELUAL.NEL.NN, I)

DIMENSION ELUAL< 1 )/ IBNCH< 7.1)

Ctttt THIS SUBROUTINE REPLACES SC BY GYRATOR AND X

Ni=IBNCH<4.I) $ N2«IBNCH<5/I)

NN=NN+1

IBNCH<2. I>=NN

IBNCH<3.I>=0

IBNCH<6.I>sl

IBfrCH<7.I>s8

ELUAL<I>=1.®

fCLsNEL+l

I8NCH< 1 , NEL )«IBNCH< 1 . I

)

I^CH<2.hEL>=Nl

IBMCH<3.hEL>=N2

I0NCM<5>ICL)«NN

IKCH<6.f€L)=l

I^CH<7,NEL)a0

ELUAL<NEL)«S.0

RETURN

END

-196-
SUBROUTINE TRLINC IBNCH.ELUAL/NEL. I.R0>

DIMENSION ELUAL< 1 ). IBNCH<7; 1

)

N1=IBNCH<4. 1 ) * N2-IBNC-HC5; I > N3=IBHCH<2. 1 ) $ N4=I^H<3.I)

is^b:h<2. I >=1H

I»CH<3.I)=iH

IBNOK6. 1 >=2

IBNCH<7.I>=0

ELiv¥^< I >=R0

^CL=^€L+l

IBNCH< 1 . ^£L >=IBNCH< 1 . 1)

IBNCH<2.h€L>=lH

IBNCH<3.^€L>=1H

IBfCH<4.h€L>=N2

iaCH<5.hEL)«Nl

IBNCH<6.^CL>=l

IBNCH<7.NEL>»3

ELUAL<KEL)=1.8

NEL=NEL+1

IBNCH< 1 . hEL)=IBNCH< 1 . 1

)

IBNCH<2.NEL)=1H

IBNCH<3.NEL>«1H

IBNCH<4.NEL>«N4

IBNCH<5.NEL>«N3

IBNCH<6/NEL>«1

IBNCH<7.NEL>«4

ELUAL<NEL>ii .8

fe.=NEL+l

ISNCHC 1 > NEL)«IBNCH< 1 . 1

)

IBNCH(2/NEL>«1H

IBNCH<3.f€L>»lH

IBNCH<4.NEL>«N3

IBNCH<5.NEL>^

IBNCH<6.h£L>2

IBNCH<7.NEL>=0

ELUAL<NEL)=R0

RETURN

END

SUBROUTINE UCCS< N1.N2>N3.N4.C0N0>G;NDIM)

DIfCNSIOH G<NOIM.n

IF <Nl.h£.0.AND.N3.NE.3) G<Nl/N3)«G<Nl.N3)-*0>©

IF CN2.Ne.0.AND.N3,h€,0) G<N2.N3>«G<N2.N3)-C0fCl

IF <Nl.NE.0.^.N4.hE.0) G<Nl.N4)«G<Nl.N4)-C0^Cl

IF <N2.NE.0.f^.N4.NE.0) G<N2.N4>«G<N2.N4>'H3>C

RETURN

END

-197-
SUBROUTIhC BNCH<N1.N2,C0ND.G.NDIM)

OlflENSION G<NOIM.. 1)

IF <Ni.h€.e> G<N1.N1)=G<N1/N1>-»C0N0

IF <N2.HE.0> G<N2..N2>=G<N2.N2HC0ND

IF <Ni.EQ.0.OR.N2.EQ.0) RETURN

G<N1.H2>=G<N1,N2)-C0ND $ G<N2.N1 >=G<N2.N1 )-aMJ

Em

SUBROUTINE SRCE<N1.N2. SOURCE. C)

DIMENSION C<1)

IF <Nl.r€.0) C<N1)«C<N1)-S0URCE

IF <N2.HE.e) C<N2>=C<N2HS0URCE

FU^CTIW NCODE<NAriE)

DIMENSION IREFC9)

DATA IREF/lHI . IHR. IHO. INC. IHN. IHU. IHL. IHT. IKE/

DO 1 1=1.9

hCODE^I

IF <N^«C.EQ.IREF<n) GO TO 2

1 CONTINUE

HCODE=10

2 RETURN

Em

-198-
REFERENCES

[I] Nahman, N.S. and Riad, S.M. , Some physical signals and systems that yield to the

homomorphic transformation, Proc. IEEE, in preparation.

[2] Oppenheim, A.V., Schafer, R.W. and Stockham, T.W., Nonlinear filtering of multiplied

and convolved signals, Proc. IEEE 56_, 1264-1291 (August 1968).

[3] Schafer, R.W., Echo removal by discrete generalized linear filtering, Research Lab.

of Electronics, MIT, Tech. Rep. 466 (1969).

[4] Oppenheim, A.V. and Schafer, R.W. , Digital Signal Processing, Chs . 2, 3, and 10

(Prentice-Hall, Englewood Cliffs, N.J., 1975).

[5] Ulrych, T.J., Application of homomorphic deconvolution to seismology. Geophysics 36 , 650-660 (August 1971)

.

[6] Nelson, R.E. and Cozyell, M.R. , Electrical parameters of precision, coaxial, airdielectric transmission lines, Nat. Bur. Stand. (U.S.), Monogr. 96 (June 1966).

[7] Riad, S.M. and Nahman, N.S., Application of the homomorphic deconvolution for the

separation of TDR signals occurring in overlapping time windows, CPEM Digest, 27-29

(June 1976) .

[8] Riad, S.M. and Nahman, N.S., Application of the homomorphic deconvolution for the

separation of IDR signals occurring in overlapping time windows, IEEE Trans. Instrun. Meas., to be published (December 1976).

[9] VonHippel, A.R. , Dielectric Materials and Applications (Chapman and Hall, London;

MIT Tech. Press and J. Wiley, 1954).

[10] VonHippel, A.R. , Dielectrics and Waves (MIT Press, Cambridge, N.Y. , 1954).

[II] Hill, N.E., et al., Dielectric Properties and Molecular Behavior (Van Nostrand Reinhold

Co., London, 1969).

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-201-
NBS-114A (REV. 7-73)

U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET

1. PL CATION OR REPORT NO.

NBSIR 78-881

2. Gov't Acces., )n No.

3. Recipient's Accession No.

4. TITLE AND SUBTITLE

"Application of the Horaomorphic Transformation

to Time Domain Measurement Problems^

5. Publication Date June 19 78

6. Performing Organization Code

724

7. AUTHOR(S)

RIAD, SEDKI M. and NAHMAN, NORRIS S.

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

Washington, D.C. 20234

10. Pro)ect/Task/ Work Unit No. 724-4153

11. Contract/Grant No. CST-8392

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

Supported in part by:

Electrical Engineering Department

University of Colorado

Boulder, Colorado

13. Type of Report & Period

Covered

14. sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most si^ificant information. If document includes a significant

bibliography or literature survey, mention it here.)

This report presents a study of the theory and application of the homomorphic

transformation to deconvolution problems occurring in time domain measurements in

the picosecond time domain. A homomorphic deconvolution transform was developed

and applied successfully to remove the time-windowing restriction required in many

time domain measurements. Examples were presented including problems in time domain

analysis of linear networks and dielectric spectroscopy, and scattering and multiple

reflection in antenna (radiation) systems were considered and treated. Also

considered was the development of a model for a 28 picosecond resolution feedthrough sampling-head, and the model's step response was computed. Simulation

studies were performed using typical input waveforms and the oscilloscope model.

The homomorphic transformation was used to deconvolve the model's impulse response

from the simulated output. Comparison of the deconvolved output waveforms with |

the input waveforms showed agreement within the accuracy of the sampled-data

simulation.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons) Antenna Scattering; deconvolut iou ; dielectric spectroscopy;

homomorphic transformation; modeling of sampling gates; signal processing;

time domain measurements; time domain ref lectometry

.

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