# Patent

## Title
DC-DC converter connected to phase locked loop

## Abstract

A DC-DC converter is provided. The DC-DC converter is connected to a PLL circuit, supplying a voltage at least to a power source terminal of a voltage controlled oscillator of the PLL circuit. A frequency of a ripple voltage included in the voltage is less than a natural frequency of the PLL circuit or more than one-half of a frequency of an output signal of the PLL circuit.

## Background
RELATED APPLICATIONS
This application claims priority to Japanese Patent Application No. 2003-416434 filed Dec. 15, 2003 which is hereby expressly incorporated by reference herein in its entirety.
BACKGROUND
1. Field of the Invention
The present invention relates to a DC-DC converter, and particularly to a DC-DC converter connected to a PLL (Phase Locked Loop) circuit.
2. Related Art
A PLL circuit has a voltage controlled oscillator to generate a signal of a desired frequency, while the voltage controlled oscillator generates a signal of the desired frequency under control of an input voltage by which the frequency of the signal is changed. The voltage controlled oscillator is supplied with a power source voltage necessary for the operation from a DC-DC converter. The DC-DC converter performs a switching operation to generate a desired DC voltage from another DC voltage, that is, a DC to DC conversion. In the voltage generated by the above-mentioned switching operation of the DC-DC converter, there is included a ripple voltage (AC component superposing on the DC voltage) resulting from the switching operation. An example of this type of DC-DC converter is described in Japanese Unexamined Patent Publication No. 2001-157437 (FIG. 12).
In the above-mentioned PLL circuit, a frequency of a signal generated by the voltage controlled oscillator fluctuates because of the ripple voltage being included in the voltage to be supplied from the DC-DC converter. Thus, a problem of generating a large amount of jitter arises.
SUMMARY
First DC-DC Converter
A first DC-DC converter according to the present invention is a DC-DC converter connected to a PLL circuit, supplying a voltage at least to a power source terminal of a voltage controlled oscillator of the PLL circuit, a frequency of a ripple voltage included in the voltage is less than a natural frequency of the PLL circuit.
According to the first DC-DC converter of the present invention, since a frequency of the ripple voltage included in the voltage to be supplied from the DC-DC converter to the PLL circuit is less than the natural frequency of the PLL circuit which is a frequency region in which a jitter value appearing in a signal generated by the PLL circuit is relatively small, the amount of the jitter value may be reduced as compared to the conventionally possible amount.
The first DC-DC converter according to the present invention performs switching for a DC to DC conversion at a frequency less than the natural frequency of the PLL circuit.
Second DC-DC Converter
A second DC-DC converter according to the present invention is a DC-DC converter connected to a PLL circuit, supplying a voltage at least to a power source terminal of a voltage controlled oscillator of the PLL circuit, a frequency of a ripple voltage included in the voltage is more than one half of the frequency of the output signal of the PLL circuit.
According to the second DC-DC converter of the present invention, since the frequency of the ripple voltage included in the voltage to be supplied from the DC-DC converter to the PLL circuit is more than one half the frequency of the output of the PLL circuit which is a frequency region in which the jitter value appearing in a signal generated by the PLL circuit is relatively small, the amount of the jitter value may be reduced as compared to the conventionally possible amount.
The second DC-DC converter according to the present invention performs switching for a DC to DC conversion at a frequency more than one half the frequency of the output of the PLL circuit.
Third DC-DC Converter
A third DC-DC converter according to the present invention is a DC-DC converter connected to a PLL circuit, supplying a voltage at least to a power source terminal of a voltage controlled oscillator of the PLL circuit, a frequency of a ripple voltage included in the voltage is an integer multiple of the frequency of the output signal of the PLL circuit.
According to the third DC-DC converter of the present invention, since the frequency of the ripple voltage included in the voltage to be supplied from the DC-DC converter to the PLL circuit is an integer multiple of the frequency of the output of the PLL circuit in which the jitter value appearing in a signal generated by the PLL circuit is virtually 0, the amount of the jitter value may be reduced as compared to the conventionally possible amount.
The third DC-DC converter according to the present invention performs switching for a DC to DC conversion at a frequency of an integer multiple of the frequency of the output signal of the PLL circuit.



BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagram showing a configuration of an oscillator of Working Example 1;
FIG. 2 is a diagram showing a configuration of a PLL circuit of Working Example 1;
FIG. 3 is a diagram showing a configuration of a DC-DC converter of Working Example 1;
FIG. 4 is a diagram of a configuration of a clock generating circuit of Working Example 1;
FIG. 5 is a graph showing a relationship between a frequency of a ripple voltage and an effective value of a period jitter of Working Example 1;
FIG. 6 is a diagram showing a configuration of a PLL circuit of Working Example 2; and
FIG. 7 is a graph showing the relationship between the frequency of the ripple voltage and the effective value of the period jitter of Working Example 2.



DETAILED DESCRIPTION
Working Example 1
FIG. 1 is a block diagram showing a configuration of an oscillator of Working Example 1 according to the present invention. As shown in FIG. 1, the oscillator of Working Example 1 consists of a PLL circuit 2 generating an oscillation clock Svco which has a desired frequency fvco based on a reference clock Sref having a reference frequency fref, and a DC-DC converter 3 generating an output voltage Vout by performing switching for a DC to DC conversion to an input voltage Vcc to supply the output voltage Vout to the PLL circuit 2.
PLL Circuit
FIG. 2 is a circuit diagram showing a configuration of a PLL circuit of Working Example 1 according to the present invention. The PLL circuit 2 of Working Example 1 has a conventionally known configuration, specifically, as shown in FIG. 2, consisting of a phase comparator 20, a charge pump 21, a loop filer 22, a voltage controlled oscillator 23, and a 1/N divider 24.
The phase comparator 20 compares a phase of the reference clock Sref which is supplied from a reference clock generator (un-illustrated) such as a quartz crystal oscillator, to a phase of a feedback clock Sfb outputted from the 1/N divider 24, specifically, comparing leading edges or trailing edges of both clocks to each other, and outputting a time difference signal Sup or Sdn, which are pulse signals showing a time difference between the both clocks, to the charge pump 21. The phase comparator 20, to be more accurate, if the feedback signal Sfb is behind the reference clock Sref, outputs the difference signal Sup indicating the amount of lag, and if the feedback signal Sfb is ahead of the reference clock Sref, outputs the difference signal Sdn indicating the amount of lead.
As shown in FIG. 2, the charge pump 21 comprises two switches, SW1 and SW2. The two switches are connected in series between the power source potential and ground potential and perform switching operation in accordance with the time difference signals Sup and Sdn. In the charger pump 21, when the time difference signal Sup is inputted from the phase comparator 20, during the time differential signal Sup is being impressed, a charging current Icp for charging the loop filter 22 is put out to the loop filter 22, while, on the other hand, when the time differential signal Sdn is inputted from the phase comparator 20, during the time difference signal Sdn is being impressed, the discharging current Icp for discharging the loop filter 22 is drawn from the loop filter 22.
The loop filter 22 comprises a resistor R and a capacitor C mutually connected in series, and by accumulating a current put out or drawn by the charger pump 21, there is generated a smoothed control voltage Vc for controlling operation of the voltage controlled oscillator 23.
The voltage controlled oscillator 23 generates an oscillation clock Svco having a frequency fvco corresponding to a level of the control voltage Vc outputted from the loop filter 22 and outputs the oscillation clock Svco to outside of the oscillator 1, while, at the same time, outputting it also to the 1/N divider 24. Specifically, the larger the control voltage Vc is, the higher the frequency of the oscillation clock Svco is, whereas, the lower the control voltage Vc is, the lower the frequency of the oscillation clock Svco is.
The 1/N divider 24 generates a feedback clock Sfb of a feedback frequency ffb by dividing the oscillation clock Svco outputted from the voltage controlled oscillator 23 by N, and outputs the feedback clock Sfb to the phase comparator 20.
As mentioned above, in the PLL circuit 2 of Working Example 1, the level of the control voltage Vc is changed to adjust the frequency fvco of the oscillation clock Svco which is generated by the voltage controlled oscillator 23 so that the phase of the reference clock Sref matches the phase of the feedback clock Sfb. As a result, when the PLL circuit 2 is operating at a steady state, a time difference or a phase difference between the reference clock Sref and the feedback clock Sfb converges to a constant steady-state error (including 0). In other words, the frequency fref of the reference clock Sref and the frequency ffb of the feedback clock Sfb become completely identical, whereupon the PLL circuit 2 stably outputs the oscillation clock Svco having a frequency fvco which is an N multiple of the frequency fref of the reference clock Sref.
DC-DC Converter and Clock Generating Circuit
FIG. 3 is a circuit diagram showing a configuration of a DC-DC converter of Working Example 1. The DC-DC converter 3 is a step-up type DC-DC converter which converts an input voltage Vcc supplied from an outside power source (un-illustrated) to an output voltage Vout of a higher voltage than the input voltage Vcc, and to be more specific, of a charge pump type DC-DC converter. The DC-DC converter 3 consists of a clock generating circuit 30 generating the reference clocks CLK1 and CLK2, n pieces (n is an arbitrary positive integer) of NMOS transistors TR1–TRn connected in series between the input voltage Vcc and the output voltage Vout, capacitors C1–Cn connected between each of NMOS transistors TR1–TRn and either the reference clocks CLK1 or CLK2, a transistor TRin inserted in a former stage of the transistor TR1, and a capacitor Cout mounted at a latter stage of the transistor TRn. The reference clocks CLK1 and CLK2 herein have a peak voltage that is the same as the input voltage Vcc and complimentary make transitions between the input potential Vcc (high level) and the ground potential (low level).
FIG. 4 is a circuit diagram showing a configuration of the clock generating circuit of Working Example 1. The clock generating circuit 30 of Working Example 1 is a ring oscillator in which k pieces (k is any arbitrary positive odd number) of inverter circuits I1–Ik are connected in a ring shape. The frequency of the reference clocks CLK1 and CLK2 generated by the clock generating circuit 30, that is the ring oscillator, are inversely proportional to the number of stages k of the inverter circuits I1–Ik and approximately inversely proportional to the square of a gate length L of MOS transistors that constitute each of the inverter circuits I1–Ik. Therefore, the frequency of the reference clocks CLK1 and CLK2 can be set at a desired frequency by varying the number of stages k and the gate length L.
Operation of the DC-DC converter will be described. To facilitate explanation and understanding, it is assumed that current consumption of the PLL circuit 2 is sufficiently small, each gate capacitance of the transistors TR1–TRn is sufficiently small as compared to the capacitance of capacitors C1–Ck, and the threshold voltage of the transistors TR1–TRn does not rise due to body effect. To further facilitate explanation, voltages of source terminals of transistors TR1–TRn are denoted as V1–Vn, and the threshold voltage of the transistors TR1–TRn is assumed to be VT.
First, when the reference clock CLK1 is at the low level, the capacitor C1 is charged by impression of the input voltage Vcc through the transistor TRin, whereby the voltage V1 becomes Vcc−VT.
Next, when the reference clock CLK1 becomes the high level and the reference clock CLK2 becomes the low level, the voltage V1 is pushed up by the reference clock CLK1 which is at the high level, to (Vcc−VT)+Vcc or 2 Vcc−VT. Thus, the voltage V2 becomes V1−VT or 2(Vcc−VT). At the same time, since the reference clock CLK2 is at the low level, a voltage Vc2 between terminals of the capacitor C2 becomes identical to the voltage V2.
Next, when the reference clock CLK2 becomes the high level, the voltage V2 is pushed up by the reference clock CLK2, which is at the high level, to 2(Vcc−VT)+Vcc or 3Vcc−VT. Thus, the voltage V3 becomes V2−VT or 3(Vcc−VT).
Thereafter, the voltage is stepped up sequentially through repetition of the above-mentioned operation. By these means, finally, the output voltage Vout is boosted up to (n+1)(Vcc−VT). In summary, the DC-DC converter 3 generates the desired output voltage Vout corresponding to the set number of stages n, thus enabling the output voltage Vout to be impressed on each composing part of the PLL circuit 2 such as the voltage controlled oscillator 23.
At this point, in the output voltage Vout, there is included a ripple voltage Vr resulting from the switching operation of the transistor TRn based on the reference clock CLK2. A relationship between the frequency fm of the ripple voltage Vr and an effective value σ of period jitter (scattering of periods) that appears in the oscillation clock Svco outputted from the voltage controlled oscillator 23 due to the ripple voltage Vr is, for example, expressed by formula (1-1),




  
    
      
        σ
        =
        
          
            
              
                K
                m
              
              ⁢
              
                V
                m
              
            
            
              
                2
              
              ⁢
              π
              ⁢
              
                
              
              ⁢
              
                f
                vco
              
            
          
          ·
          
            
              f
              m
            
            
              
                
                  
                    (
                    
                      
                        f
                        m
                        2
                      
                      -
                      
                        f
                        n
                        2
                      
                    
                    )
                  
                  2
                
                +
                
                  
                    (
                    
                      2
                      ⁢
                      
                        
                      
                      ⁢
                      ζ
                      ⁢
                      
                        
                      
                      ⁢
                      
                        f
                        m
                      
                      ⁢
                      
                        f
                        n
                      
                    
                    )
                  
                  2
                
              
            
          
          ·
          
            
            
              sin
              ⁢
              
                
              
              ⁢
              
                (
                
                  
                    
                      f
                      m
                    
                    
                      f
                      vco
                    
                  
                  ⁢
                  π
                
                )
              
            
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          1
        
        )
      
    
  




where Km (unit: Hz/V) is a fluctuation sensitivity of an oscillation frequency of the voltage controlled oscillator relative to the ripple voltage, Vm (unit: V) is an amplitude of the ripple voltage, fm (unit: Hz) is a frequency of the ripple voltage, fn (unit: Hz) is a natural frequency of the PLL circuit 2, and ζ is a dumping coefficient of the PLL circuit 2.
Derivation of the Period Jitter Equation
Description will be made regarding the derivation of the equation (1-1) which shows the relationship between the ripple voltage Vr included in the output voltage Vout, which is outputted from the DC-DC converter 3 to the voltage controlled oscillator 23, and the effective value σ of the period jitter of the oscillation clock Svco generated by the voltage controlled oscillator 23.
An amount of phase fluctuation (effective value) of the oscillation clock Svco generated by the ripple voltage Vr is expressed by formula (1–2),




  
    
      
        
          Δ
          ⁢
          
            
          
          ⁢
          
            φ
            
              R
              ⁢
              
                
              
              ⁢
              M
              ⁢
              
                
              
              ⁢
              S
            
          
        
        =
        
          
            
              G
              ⁡
              
                (
                
                  f
                  m
                
                )
              
            
            
              2
            
          
          ⁢
          
            (
            
              
                
                  K
                  m
                
                ⁢
                
                  V
                  m
                
              
              
                f
                m
              
            
            )
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          2
        
        )
      
    
  




where G(fm) is a function representing a degree of suppressing phase fluctuation due to phase locking operation of the PLL circuit 2.
Employing the formula (1-2), single side band power spectrum Sφ(f) of the phase fluctuation appearing in the oscillation clock Svco is expressed by formula (1-3),




  
    
      
        
          
            S
            φ
          
          ⁡
          
            (
            f
            )
          
        
        =
        
          
            Δ
            ⁢
            
              
            
            ⁢
            
              φ
              
                R
                ⁢
                
                  
                
                ⁢
                M
                ⁢
                
                  
                
                ⁢
                S
              
            
            ⁢
            
              
                
                   
                  2
                
                ⁢
                δ
              
              ⁡
              
                (
                
                  
                    f
                    m
                  
                  -
                  f
                
                )
              
            
          
          =
          
            
              
                
                  G
                  ⁡
                  
                    (
                    
                      f
                      m
                    
                    )
                  
                
                2
              
              2
            
            ⁢
            
              
                (
                
                  
                    
                      K
                      m
                    
                    ⁢
                    
                      V
                      m
                    
                  
                  
                    f
                    m
                  
                
                )
              
              2
            
            ⁢
            
              δ
              ⁡
              
                (
                
                  
                    f
                    m
                  
                  -
                  f
                
                )
              
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          3
        
        )
      
    
  




where δ(•) is the Dirac delta function.
Once the single side band power spectrum Sφ(f) of the phase fluctuation is defined, the effective value σ of the period jitter is given by formula (1-4).




  
    
      
        
          
            
              σ
              =
              
              ⁢
              
                
                  1
                  
                    π
                    ⁢
                    
                      
                    
                    ⁢
                    
                      f
                      vco
                    
                  
                
                ⁢
                
                  
                    
                      ∫
                      0
                      ∞
                    
                    ⁢
                    
                      
                        
                          S
                          φ
                        
                        ⁡
                        
                          (
                          f
                          )
                        
                      
                      ⁢
                      
                        
                          sin
                          2
                        
                        ⁡
                        
                          (
                          
                            
                              f
                              
                                f
                                vco
                              
                            
                            ⁢
                            π
                          
                          )
                        
                      
                      ⁢
                      
                        
                      
                      ⁢
                      
                        d
                        f
                      
                    
                  
                
              
            
          
        
        
          
            
              =
              
              ⁢
              
                
                  G
                  ⁡
                  
                    (
                    
                      f
                      m
                    
                    )
                  
                
                ⁢
                
                  
                    
                      K
                      m
                    
                    ⁢
                    
                      V
                      m
                    
                  
                  
                    
                      2
                    
                    ⁢
                    π
                    ⁢
                    
                      
                    
                    ⁢
                    
                      f
                      vco
                    
                    ⁢
                    
                      f
                      m
                    
                  
                
                ⁢
                
                  
                  
                    sin
                    ⁢
                    
                      
                    
                    ⁢
                    
                      (
                      
                        
                          
                            f
                            m
                          
                          
                            f
                            vco
                          
                        
                        ⁢
                        π
                      
                      )
                    
                  
                  
                
              
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          4
        
        )
      
    
  




If the PLL circuit 2 is a system having the simplest second order transfer function as illustrated in FIG. 2, the function G(fm) in the formula (1-2) shows a typical second order low-frequency cut-off characteristic and is expressed by formula (1-5).




  
    
      
        
          G
          ⁡
          
            (
            
              f
              m
            
            )
          
        
        =
        
          
            f
            m
            2
          
          
            
              
                
                  (
                  
                    
                      f
                      m
                      2
                    
                    -
                    
                      f
                      n
                      2
                    
                  
                  )
                
                2
              
              +
              
                
                  (
                  
                    2
                    ⁢
                    
                      
                    
                    ⁢
                    ζ
                    ⁢
                    
                      
                    
                    ⁢
                    
                      f
                      m
                    
                    ⁢
                    
                      f
                      n
                    
                  
                  )
                
                2
              
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          5
        
        )
      
    
  




The natural frequency fn and the dumping coefficient ζ of the PLL circuit 2 are given by formula (1-6) and formula (1-7), respectively,




  
    
      
        
          f
          n
        
        =
        
          
            1
            
              2
              ⁢
              
                
              
              ⁢
              π
            
          
          ⁢
          
            
              
                
                  I
                  cp
                
                CN
              
              ⁢
              
                K
                v
              
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          6
        
        )
      
    
  
  
    
      
        
          ζ
          =
          
            
              R
              2
            
            ⁢
            
              
                
                  
                    
                      I
                      cp
                    
                    ⁢
                    C
                  
                  N
                
                ⁢
                
                  K
                  v
                
              
            
          
        
        ⁢
        
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          7
        
        )
      
    
  




where Kv (unit: Hz/V) represents sensitivity of the voltage controlled oscillator 23 relative to the control voltage Vc.
If the effective value σ of the period jitter shown by the formula (1-4) is rewritten, it may result in the formula (1-1).




  
    
      
        σ
        =
        
          
            
              
                K
                m
              
              ⁢
              
                V
                m
              
            
            
              
                2
              
              ⁢
              π
              ⁢
              
                
              
              ⁢
              
                f
                vco
              
            
          
          ·
          
            
              f
              m
            
            
              
                
                  
                    (
                    
                      
                        f
                        m
                        2
                      
                      -
                      
                        f
                        n
                        2
                      
                    
                    )
                  
                  2
                
                +
                
                  
                    (
                    
                      2
                      ⁢
                      
                        
                      
                      ⁢
                      ζ
                      ⁢
                      
                        
                      
                      ⁢
                      
                        f
                        m
                      
                      ⁢
                      
                        f
                        n
                      
                    
                    )
                  
                  2
                
              
            
          
          ·
          
            
            
              sin
              ⁢
              
                
              
              ⁢
              
                (
                
                  
                    
                      f
                      m
                    
                    
                      f
                      vco
                    
                  
                  ⁢
                  π
                
                )
              
            
            
          
        
      
    
    
      
        (
        
          1
          ⁢
          
            -
          
          ⁢
          1
        
        )
      
    
  




Graph of Periodic Jitter
FIG. 5 is a graph showing a relationship between the frequency fm of the ripple voltage and the effective value σ of the period jitter defined in the formula (1-1). The graph of FIG. 5 is drawn under the condition that Vm=10 mV, Km=10 MHz/V, fvco=100 MHz, and ζ=0.7.
As apparent from the graph of FIG. 5, the effective value σ of the period jitter is large when the frequency fm of the ripple voltage Vr is in a range approximately from the natural frequency fn of the PLL circuit 2 to 1⁄2 of the frequency fvco of the oscillation clock Svco of the PLL circuit 2. On the contrary, it is small when the frequency fm of the ripple voltage Vr is below the natural frequency fn of the PLL circuit 2 or over 1⁄2 of the frequency fvco of the oscillation clock Svco of the PLL circuit 2. Further, as likewise apparent from the graph of FIG. 5, the effective value σ of the period jitter is 0 when the frequency fm of the ripple voltage Vr is an integer multiple of the frequency fvco of the oscillation clock Svco.
In the oscillator 1 illustrated in FIG. 1, according to the results of consideration based on the graph of FIG. 5 as described above, the clock generating circuit 30 shown in FIG. 4, which is provided on the DC-DC converter 3 illustrated in FIG. 3, outputs the reference clocks CLK1 and CLK2 having a frequency corresponding to any of (1) less than the natural frequency fn of the PLL circuit 2, (2) over 1⁄2 of the frequency fvco of the oscillation clock Svco of the PLL circuit 2, or (3) the integer multiple of the oscillation clock Svco. Since the DC-DC converter 3 outputs, to the voltage controlled oscillator 23, the output voltage Vout generated by the switching operation based on the reference clocks CLK1 and CLK2, there is no superposition of a ripple voltage Vr which may have a frequency component functioning to increase the period jitter of the oscillation clock Svco of the voltage controlled oscillator 23. Therefore, the amount of the period jitter of the oscillation clock Svco, which is outputted by the voltage controlled oscillator 23 operated by the output voltage Vout, can be reduced as apparent from the graph of FIG. 5.
Working Example 2
An oscillator of Working Example 2 will be described.
FIG. 6 is a circuit diagram showing a configuration of a PLL circuit of Working Example 2. The oscillator of Working Example 2 has the same configuration as the oscillator 1 of Working Example 1 illustrated in FIG. 1, and to be more accurate, having the PLL circuit 4 shown in FIG. 6 instead of the PLL circuit 2 shown in FIG. 2 as well as the DC-DC converter 3 shown in FIG. 3.
As shown in FIG. 6, the PLL circuit 4 comprises the phase comparator 40, the charge pump 41, the loop filer 42, the voltage controlled oscillator 43, and the 1/N divider 44, which respectively have the same configuration and function as the phase comparator 20, the charge pump 21, the loop filer 22, the voltage controlled oscillator 23, and the 1/N divider 24 in the PLL circuit 2 of Working Example 1 shown in FIG. 2, and additionally including an 1/X divider 45 outputting a dividing clock Sdiv having a frequency fdiv corresponding to 1/X of that of the oscillation clock Svco.
An effective value σ of period jitter that appears in the dividing clock Sdiv outputted from the 1/X divider 45 is expressed by formula (2-1).




  
    
      
        
          
            
              σ
              =
              
              ⁢
              
                
                  1
                  
                    π
                    ⁢
                    
                      
                    
                    ⁢
                    
                      f
                      div
                    
                  
                
                ⁢
                
                  
                    
                      ∫
                      0
                      ∞
                    
                    ⁢
                    
                      
                        
                          
                            S
                            φ
                          
                          ⁡
                          
                            (
                            f
                            )
                          
                        
                        
                          X
                          2
                        
                      
                      ⁢
                      
                        
                          sin
                          2
                        
                        ⁡
                        
                          (
                          
                            
                              f
                              
                                f
                                div
                              
                            
                            ⁢
                            π
                          
                          )
                        
                      
                      ⁢
                      
                        
                      
                      ⁢
                      
                        d
                        f
                      
                    
                  
                
              
            
          
        
        
          
            
              =
              
              ⁢
              
                
                  
                    
                      G
                      ⁡
                      
                        (
                        
                          f
                          m
                        
                        )
                      
                    
                    X
                  
                  ·
                  
                    
                      
                        K
                        m
                      
                      ⁢
                      
                        V
                        m
                      
                    
                    
                      
                        2
                      
                      ⁢
                      π
                      ⁢
                      
                        
                      
                      ⁢
                      
                        f
                        div
                      
                      ⁢
                      
                        f
                        m
                      
                    
                  
                
                ⁢
                
                  
                  
                    sin
                    ⁢
                    
                      
                    
                    ⁢
                    
                      (
                      
                        
                          
                            f
                            m
                          
                          
                            f
                            div
                          
                        
                        ⁢
                        π
                      
                      )
                    
                  
                  
                
              
            
          
        
      
    
    
      
        (
        
          2
          ⁢
          
            -
          
          ⁢
          1
        
        )
      
    
  




By rewriting in the same manner as rewriting formula (1-4) to the formula (1-1) in Working Example 1, the formula (2-1) may be modified as formula (2-2).




  
    
      
        σ
        =
        
          
            
              
                K
                m
              
              ⁢
              
                V
                m
              
            
            
              
                2
              
              ⁢
              π
              ⁢
              
                
              
              ⁢
              
                f
                vco
              
            
          
          ·
          
            
              f
              m
            
            
              
                
                  
                    (
                    
                      
                        f
                        m
                        2
                      
                      -
                      
                        f
                        n
                        2
                      
                    
                    )
                  
                  2
                
                +
                
                  
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FIG. 7 is a graph showing a relationship between the frequency fm of the ripple voltage Vr and the effective value σ of the period jitter. The graph of FIG. 7 is drawn under the condition that X=10 and other conditions are the same as those of the graph of FIG. 5.
As apparent from the graph of FIG. 7, in the same way as the graph of FIG. 5 of Working Example 1, the effective value σ of the period jitter is small when the frequency fm of the ripple voltage Vr is approximately below the natural frequency fn of the PLL circuit 4 and over 1⁄2 of the frequency fdiv of the dividing clock Sdiv of the PLL circuit 4, and 0 when the frequency fm of the ripple voltage Vr is an integer multiple of the frequency fdiv of the dividing clock Sdiv of the PLL circuit 4.
According to the results of consideration based on the graph of FIG. 7 as described above, in the same way as the oscillator 1 of Working Example 1, also in the oscillator of Working Example 2, the clock generating circuit 30 outputs the reference clocks CLK1 and CLK2 having a frequency corresponding to any of (1) less than the natural frequency fn of the PLL circuit 4, (2) over 1⁄2 of the frequency fdiv of the dividing clock Sdiv of the PLL circuit 4, or (3) the integer multiple of the dividing clock Sdiv. Since the DC-DC converter 3 outputs, to the voltage controlled oscillator 43, the output voltage Vout generated by the switching operations based on the reference clocks CLK1 and CLK2, there is no superposition of a ripple voltage Vr which may have a frequency component functioning to increase the period jitter of the dividing clock Sdiv of the voltage controlled oscillator 43. Therefore, the amount of the period jitter of the dividing clock Sdiv, which is outputted by the voltage controlled oscillator 43 operated by the output voltage Vout, can be reduced as apparent from the graph of FIG. 7.
It should be noted that in the two working examples described above, description was made by using the step-up DC-DC converter of the charge pump type, whereas, be it a chopper type DC-DC converter or a step-down DC-DC converter, the same means may produce the same results.

## Claims

1. A DC-DC converter connected to a PLL circuit comprising:
a clock generating circuit that generates first and second reference clock signals;
a plurality of transistors connected in series that generate an output voltage based on the first and second reference clock signals and an input voltage, the output voltage powers a voltage controlled oscillator of the PLL circuit and the voltage controlled oscillator generates an oscillation clock based on a control voltage; and
a plurality of capacitors, each of said plurality of capacitors corresponds to each of the plurality of transistors and connects the each of the plurality of transistors to one of the first reference signal and the second reference signal, wherein
the output voltage includes a ripple voltage generated based on switching one of the plurality of transistors, the ripple voltage having a frequency less than a natural frequency of the PLL circuit.

2. The DC-DC converter according to claim 1, wherein the DC-DC converter performs a switching operation for a DC to DC conversion at a frequency of less than the natural frequency of the PLL circuit.

3. A DC-DC converter connected to a PLL circuit comprising:
a clock generating circuit that generates first and second reference clock signals;
a plurality of transistors connected in series that generate an output voltage based on the first and second reference clock signals and an input voltage, the output voltage powers a voltage controlled oscillator of the PLL circuit and the voltage controlled oscillator generates an oscillation clock based on a control voltage; and
a plurality of capacitors, each of said plurality of capacitors corresponds to each of the plurality of transistors and connects the each of the plurality of transistors to one of the first reference signal and the second reference signal, wherein
the output voltage includes a ripple voltage generated based on switching one of the plurality of transistors, the ripple voltage having a frequency more than one-half of a frequency of an output signal of the PLL circuit.

4. The DC-DC converter according to claim 3, wherein the DC-DC converter performs a switching operation for a DC to DC conversion at a frequency of more than one-half of the frequency of the output signal of the PLL circuit.

5. A DC-DC converter connected to a PLL circuit comprising:
a clock generating circuit that generates first and second reference clock signals;
a plurality of transistors connected in series that generate an output voltage based on the first and second reference clock signals and an input voltage, the output voltage powers a voltage controlled oscillator of the PLL circuit and the voltage controlled oscillator generates an oscillation clock based on a control voltage; and
a plurality of capacitors, each of said plurality of capacitors corresponds to each of the plurality of transistors and connects the each of the plurality of transistors to one of the first reference signal and the second reference signal, wherein
the output voltage includes a ripple voltage generated based on switching one of the plurality of transistors, the ripple voltage having a frequency that is an integer multiple of a frequency of the output signal of the PLL circuit.

6. The DC-DC converter according to claim 5, wherein the DC-DC converter performs a switching for a DC to DC conversion at a frequency which is an integer multiple of the frequency of the output signal of the PLL circuit.