# Patent

## Title
Integrated circuit dynamic parameter management in response to dynamic energy evaluation

## Abstract

A single integrated circuit (12). The integrated circuit comprises a first circuit (14x) having a data path, the first circuit consisting of a first number of logic gates for performing a plurality of logic functions. The integrated circuit also comprises a circuit (22x) for indicating a potential speed capability of the data path. The circuit for indicating comprises a second number of logic gates (82, 92) for performing the plurality of logic functions, wherein the second number is less than the first number. The circuit for indicating also comprises additional circuitry (88, 98) for representing parasitic characteristics in the data path.

## Background
CROSS-REFERENCES TO RELATED APPLICATIONS
This divisional application claims priority, under 35 U.S.C. §119(e)(1), of U.S. patent application Ser. No. 10/739,787, filed on Dec. 18, 2003 now U.S. Pat. No. 7,131,089 which was filed as a non-provisional application of U.S. Provisional Application No. 60/480,053, filed Jun. 20, 2003 and 60/480,151, filed Jun. 20, 2003, and incorporated herein by this reference.


This application relates to U.S. patent application Ser. No. 10/739,469 (TI-36581), entitled “Integrated Circuit Dynamic Parameter Management In Response To Dynamic Energy Evaluation”, and filed on the same date as the present application.
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT
Not Applicable.
BACKGROUND OF THE INVENTION
The present embodiments relate to electronic circuits and are more particularly directed to an integrated circuit speed capability indicator that may be used for various purposes including performance control and energy efficiency adjustments for an integrated circuit.
Electronic circuits are prevalent in numerous applications and are used in devices in personal, business, and other environments. Demands of the marketplace affect many design aspects of these circuits, including device size, complexity, efficiency, and cost. These aspects are often important in various devices. By way of example, the mobile phone industry is transitioning from devices that are voice oriented to devices that are multimedia oriented, and multimedia applications typically integrate high performance processing cores. As a result, the above-mentioned aspects of size, complexity, efficiency, and cost manifest themselves in various areas, including energy consumption and battery lifetime. These areas are also a concern in various other electronic devices, particularly where energy is a concern such as in other battery-powered applications. Thus, to maintain pace with marketplace demands and supplier goals for these devices, considerations with respect to these factors are of paramount interest. The preferred embodiments are directed to these aspects.
Along with the proliferation of various electronic devices there is also an ongoing effort in the electronics industry to continue to reduce individual component sizes. This latter effort is sometimes described as sub-micron technology, when describing that the gate length of a single transistor on the electronic device is less than one micron. Of course, because a common electronic device is formed with an increasingly large number of transistors, then by reducing the length of each such device, the overall electronic device is reduced in size. Indeed, transistor lengths are now reducing in size toward an area sometimes referred to as ultra-deep sub-micron, corresponding to an ongoing trend to further reduce the size of transistor gate lengths. For example, current technology contemplates transistor gate lengths on the order of 90 nanometers, or even a lesser amount of 65 nanometers, with the expectation that even further size reductions are contemplated.
With the developments described above, the current state of the art uses two known techniques in an effort to control power consumption on certain devices. In a first technique, at certain times of operation, the global power supply voltage and frequency to the device are reduced. In a second technique, the threshold voltage of the transistors on the device is increased, such as by adjusting a back bias to each of the transistors. While each of these approaches has been used in the past to control power consumption, they have certain drawbacks that have been discovered by the present inventors in connection with the preferred embodiments, particularly as transistor geometries continue to decrease. By way of example, it has been observed that the first technique, reduction of the global power supply voltage, may actually cause an increase in total loss of energy per cycle when leakage current is taken into account. More particularly, it is known that for an overall integrated circuit and particularly in contemporary design, at a given time only a portion of the total circuit is operational, while the remaining transistors in the device remain at their respective non-switched states. This is increasingly the case for system-on-chip (“SOC”) devices, where a number of different functional circuits are formed on a single integrated circuit die. On the SOC or a comparable device, the transistors that are temporarily not switched may leak current at a same time that the then-operating transistors are using functionally necessary current to switch state. The present inventors have observed that, for various reasons, the leakage of these non-switched transistors grows ever more considerable relative to the overall device as device geometries decrease. For example, with the reduction in transistor size, the transistor gate oxide, that is, the insulator between the transistor gate and the underlying semiconductor region, is thinner and, thus, current is more likely to leak through this thinner region. As another example, due to the SOC nature of a device as well as the goal of providing high speed switching devices, there is a lesser ability to temporarily isolate the non-switching transistors so as to avoid or reduce the effect of their current leakage. In any event, therefore, when total leakage current is taken into account for the SOC or like device, then lowering the global power supply voltage and as frequency decreases will actually cause more power consumption than if the supply voltage were left alone or as compared to power consumption at higher frequencies. Also in connection with smaller device geometries, the present inventors have observed that increasing transistor threshold voltages by adjusting back bias will moderately reduce power consumption in some instances, yet the amount of reduction may be further improved upon as is achieved with the preferred embodiments.
In view of the above, there is a need to further improve aspects relating to circuit and device performance control and energy efficiency. In one sense, these aspects may be evaluated and potentially responded to by providing an indicator that predicts or provides a guideline to the anticipated operational capability of a larger circuit corresponding to that indicator. The preferred embodiments are directed to such an indicator, and further include an overall system in which the indicator may be included, thereby addressing the numerous aspects discussed above and improving upon the prior art.
BRIEF SUMMARY OF THE INVENTION
In one preferred embodiment, there is a single integrated circuit. The integrated circuit comprises a first circuit having a data path, the first circuit consisting of a first number of logic gates for performing a plurality of logic functions. The integrated circuit also comprises a circuit for indicating a potential speed capability of the data path. The circuit for indicating comprises a second number of logic gates, wherein the second number is less than the first number and provides voltage and delay characteristics comparable to the data path. The circuit for indicating also comprises additional circuitry for representing parasitic characteristics in the data path.
In another preferred embodiment, there is a computer programmed to specify a design of a circuit for indicating a potential speed capability of a data path in a predetermined circuit. The data path comprises a plurality of logic functions to be performed by a first number of logic gates. The computer specifies the design by steps. In one step, the program specifies a second number of logic gates to be included in the circuit for indicating a potential speed capability. The second number of logic gates is less than the first number of logic gates and provides voltage and delay characteristics comparable to the data path. The computer also specifies additional circuitry to be included in the circuit for indicating a potential speed capability, wherein the additional circuitry is for representing parasitic characteristics in the data path.
Other aspects are also disclosed and claimed.



BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING
FIG. 1 illustrates a functional block diagram of a system 10 according to a preferred embodiment.
FIG. 2a illustrates a simplified block diagram of certain components that may be used to provide the functionality of either leakage monitor 241, or 242 of FIG. 1.
FIG. 2b illustrates a timing diagram in connection with the operation of the block diagram of FIG. 2a. 
FIG. 3 illustrates a functional block diagram of controller 26 of FIG. 1.
FIG. 4 illustrates a state diagram of an example of operation of controller 26 of FIG. 3.
FIG. 5 illustrates a block diagram of a general architecture that may be used to form speed each capability indicator 22x of FIG. 1 as a ring oscillator.
FIG. 6 illustrates a graph having voltage along its horizontal axis and delay along its vertical axis, where the graph depicts five plots to demonstrate different voltage/delay characteristics of five different respective data paths from which a critical path may be selected.
FIG. 7 illustrates a schematic of a preferred implementation for each blocks 22xA, 22xB, and 22xC in FIG. 5.
FIG. 8a illustrates a graph having voltage along its horizontal axis and frequency along its vertical axis, and it further illustrates a plot CPP as the voltage/frequency characteristics of a critical path CPx and a plot PF1 as the voltage/frequency characteristics of a speed capability indicator 22x that is attempting to match the characteristics of that critical path CPx.
FIG. 8b illustrates the graph of FIG. 8a after adjustments are made to plot PF1 so as to better match it to the voltage/frequency characteristics of the corresponding critical path CPx.



DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 illustrates a functional block diagram of a system 10 according to a preferred embodiment. System 10 may represent any of various different types of electronic devices, where those devices provide functionality through the use of one or more integrated circuit devices. Indeed, as indicated in the preceding Background Of The Invention section of this document, the preferred embodiments have particular benefit in devices where power and energy consumption are considerations, such as in mobile phones and other battery-operated devices, so system 10 may represent a portion of any one of those devices. In a general sense, various aspects of system 10 are known in the art, as are variations that may be ascertained by one skilled in the art, where those variations are also contemplated as within the present inventive scope. Thus, the following first describes these known aspects by way of introduction, followed by a detailed discussion of aspects which in combination with the known aspects provide for the inventive preferred embodiments.
System 10 is shown by example to include one system-on-a-chip (“SOC”) block 12, which represents a single integrated circuit device that encompasses various functionality. With respect to that functionality, and by way of example, SOC block 12 is shown to include two different cores 141 and 142, each representing an example of certain major functions associated with SOC block 12. For example, core 141 could be a general purpose microprocessor core, while core 142 could be a digital signal processor (“DSP”) core; however, alternative functioning blocks as well as a different number of functional blocks could be implemented and benefit from the inventive teachings of this document. In addition to cores 141 and 142, SOC block 12 may include additional functionality, such as supportive or ancillary functionality with respect to cores 141 and 142. As an illustration in this regard, SOC block 12 is also shown to include two functional blocks 161 and 162, where the choice of two such blocks is solely by way of example.
Each functioning circuit of SOC block 12, including cores 141 and 142 and functional blocks 161 and 162, is connected to receive a power supply voltage from a power supply 18, through a power bus PB. In one approach, the same power supply voltage is provided to all of these circuits via power bus PB, and in an alternative approach power bus PB may have multiple conductors whereby different voltages may be supplied to different circuits within SOC block 12. Power bus PB is intended to be connected in a manner to provide, from power supply 18, a general system bias voltage, VDD, to the circuits to which it is connected. As known in the art, therefore, the system bias voltage, VDD, may be connected to various devices, such as selected source, drain, and/or gate nodes of various transistors within system 10. In addition, however, and for reasons more evident below, power bus PB is also intended to be connected in a manner to provide, from power supply 18, a threshold voltage altering system back bias voltage, VBB, to the back gates or other appropriate threshold voltage connections of at least some of the transistors in the various circuits to which power bus PB is connected. Further in this regard, while a single back bias voltage, VBB, is shown, in the preferred embodiment two separate back bias voltages are provided, one for p-channel transistors and one for n-channel transistors, where as known in the art a different amount of voltage adjustment is required to cause an equal change in the threshold voltage of these different conductivity type devices. Note also that in the example of FIG. 1, power supply 18 is shown external from SOC block 12 as is common in contemporary applications. However, the preferred embodiments also contemplate that when feasible as may be ascertainable by one skilled in the art, power supply 18 may be integrated within the same single integrated circuit that forms SOC block 12. Lastly, for sake of simplicity, power supply 18 is only shown as providing one system value of VDD and one system value of VBB. However, the preferred embodiments also contemplate, when feasible, providing different sets of VDD and VBB, such as one set of VDD and VBB to core 141 and a separately adjustable and, thus, potentially different, set of VDD and VBB to core 142.
Each functioning circuit of SOC block 12, including cores 141 and 142 and functional blocks 161 and 162, is connected to receive a system clock signal, via a clock bus CB, from a system clock 20. In general, the frequency of the system clock signal governs the speed of operation of the circuits receiving that signal. System clock 20 also receives a voltage supply from power supply 18. Also, as known in the art, the frequency of the clock signal provided by system clock 20 to clock bus CB is changed from time to time, typically under the control of the system operating system. Lastly, note by way of example that system clock 20 is shown to be external from SOC block 12, but in an alternative embodiment it may be integral within SOC block 12.
Turning now to various aspects of SOC block 12 as distinguishable from the prior art, in the preferred embodiment each core 14x includes two additional corresponding circuits, where each of those circuits is now introduced and as shown later where the combination of those circuits provides for enhanced performance control of system 10 in the form of energy savings and/or processing capability. Looking then to the two additional circuits in core 141 by way of example, they include a speed capability indicator 221 and a leakage monitor 241. Additionally, having introduced speed capability indicator 221 and leakage monitor 241, note that core 142 includes comparable devices, namely, a speed capability indicator 222 and a leakage monitor 242. These devices perform the same functions, relative to core 142, as described with respect to each counterpart in core 141. Further, SOC block 12 includes a controller 26, which is shared in the sense of signal evaluation and functionality between cores 141 and 142. Lastly, note that with certain variations as ascertainable by one skilled in the art, in alternative embodiments a different number of speed capability indicators and/or leakage monitors may be implemented as within the present inventive scope.
Looking in detail to speed capability indicator 221, it is constructed and operates in response to system voltages VDD and VBB to provide resultant voltage and speed characteristics that in general mimic the voltage and speed characteristics of the critical path in core 141. More particularly, a critical path is identifiable in a core as is known in the art, such as through the use of various diagnostic software during the development of the core design. Typically, a critical path provides the most restrictive bottleneck to the data processing of a given circuit. For sake of illustration, such a critical path for core 141 is shown generally as a critical path CP1. In the preferred embodiment, with that core critical path CP1 identified, speed capability indicator 221 is designed and formed as a circuit that is preferably much smaller than critical path CP1, where this smaller circuit that forms speed capability indicator 221 operates in response to the same system voltages VDD and VBB as critical path CP1, but where speed capability indicator 221 is designed to repeatedly switch state at a rate that is responsive to those two system voltages. This switching rate, or frequency, or a signal responsive to it, is provided by an output signal that is a measure of that switching rate, where in one preferred embodiment the switching rate is output directly as an output frequency, fS1; in alternative embodiments, however, different measures of the switching rate may be provided, such as by outputting a digital count of or proportional to the switching rate. In any event, in the preferred embodiment wherein fS1 is a frequency signal, the switching rate of fS1 is governed by a feedback control in speed capability indicator 221 and not the system clock from clock bus CB. Thus, variations in VDD and/or VBB can affect fS1, while fS1 is not affected by the system clock from clock bus CB. Nonetheless, according to one preferred embodiment, and due to the attempt to match the characteristics of speed capability indicator 221 with those of core critical path CP1, then typically it is anticipated that the frequency capability of speed capability indicator 221 will mimic that of critical path CP1 of core 141, that is, for a given system voltage VDD and a given system back bias voltage VBB, critical path CP1 will be expected to operate at a frequency that is close to, or the same as, the switching frequency, fS1, of speed capability indicator 221. Further, in an alternative preferred embodiment, switching frequency, fS1, gives a downward-scaled indication of critical path speed, so as to reduce power consumption of that indicator. For example, in one preferred embodiment, an explicit frequency divider may be included in speed capability indicator 221 so as to reduce the resulting switching frequency, fS1, which by way of example may be reduced to represent a speed that is 50% of that expected of critical path CP1.
In the preferred embodiment, various aspects are implemented in an effort to ensure the likeness of voltage/frequency characteristics as between core critical path CP1 and speed capability indicator 221. As one aspect, in the preferred embodiment various design software, as available in the art, is operable to receive the network node listings or the like of critical path CP1 and to provide a physically smaller design for speed capability indicator 221, that is, for a smaller circuit that provides comparable voltage and frequency response to that of critical path CP1. With respect to the size difference between critical path CP1 and speed capability indicator 221, for 65 nm technology, critical path CP1 may be on the order of 500 μm in one dimension along an integrated circuit while speed capability indicator 221 is by way of comparison on the order of 50 μm in one dimension along an integrated circuit; speed capability indicator 221 is therefore less than or equal to approximately ten percent in size relative to critical path CP1. Also using such software, in the preferred embodiment various iterations are repeated at the design stage so as to fine tune the voltage and frequency characteristics of speed capability indicator 221 to approach those of critical path CP1 in core 141, or scaled values thereof. In addition, however, in the preferred embodiment speed capability indicator 221 is a tunable circuit, that is, provisions are made so that after it is implemented as a semiconductor device, additional tuning (or “trimming”) may be made to the circuit so as to further refine its voltage/frequency characteristics to duplicate those of critical path CP1 in core 141. This ability to tune may be implemented in various manners as ascertainable by one skilled in the art, and indeed may be done in a manner such that digital tuning values can be loaded into speed capability indicator 221 after fabrication so as to accomplish a high degree of precision in the ability to tune the circuit as well as to provide simplicity and compactness as compared to other methods of trimming (e.g., tunable capacitors). As another aspect, speed capability indicator 221 is located proximate the corresponding critical path CP1 in core 141. For example, for 90 nanometer technology, then the distance between speed capability indicator 221 and critical path CP1 is preferably less than 100 μm. As another aspect, in the preferred embodiment speed capability indicator 221 is constructed to be approximately square in layout along the semiconductor device that forms SOC block 12. Note that the proximity and layout preferences are implemented in an effort to have the same factors, such as process and operational factors, comparably influence both critical path CP1 in core 141 and speed capability indicator 221. For example, process variations, which cause peaks and valleys in operational aspects in both of the two dimensions laterally across the semiconductor device, will preferably influence both critical path CP1 in core 141 and speed capability indicator 221 in a comparable manner. As another example, temperature variations will preferably influence both critical path CP1 in core 141 and speed capability indicator 221 in a comparable manner. In any event, therefore, once speed capability indicator 221 is properly tuned, it is thereafter anticipated that in ideal conditions it will switch state, as reflected in fS1, at a rate that is the same as (or proportional to) the rate of operation for the corresponding core 141. However, under non-ideal conditions, and as detailed later, the corresponding core 141 may operate at a different rate (taking into any scaling, if applicable) than reflected in fS1, and this difference forms a basis in the preferred embodiments by which performance may be adjusted to improve overall efficiency. From the preceding considerations, therefore, one skilled in the art may readily ascertain various manners to design and fabricate speed capability indicator 221. In addition to those ascertainable approaches, still additional considerations toward preferred embodiments for speed capability indicator 221 are described later, with respect to FIGS. 5 through 8b. 
Looking in detail to leakage monitor 241, it is constructed and operates to provide a signal indication, fL1, representing the amount of current leakage in the corresponding core 141; by way of preferred examples, this indication may be a signal, such as a signal having a frequency or a count, that is either proportional to the leakage directly or that provides a ratio of the leakage current relative to the active current. Such a leakage circuit may be designed by one skilled in the art. By way of example and as guidance in this endeavor, FIG. 2a illustrates a simplified example of a block diagram of certain components that may be used to provide such functionality, and since that block diagram may represent either leakage monitor 241 or 242, it is shown generally as a leakage monitor 24x. Leakage monitor 24x includes an active and leakage mimic block 100, which is so named because it includes circuitry sufficient to mimic the active and leakage behavior of the corresponding core 14x that monitor 24x seeks to represent in terms of a leakage current indication. Note that the specific design for mimic block 100 in the preferred embodiment is preferably integrated into some part of the automated design of the corresponding circuit core, which in this regard is comparable to that of the design of a speed capability indicator 22x. In other words, for a given design and the software that produces it, that same software may be modified so as to produce, such as from the network list of the core design, a comparable design for a simplified circuit that produces leakage consistent with the corresponding core, where that circuit thus forms mimic block 100. Thus, for all integrated circuits sharing a same design, each will also share a comparable design of a corresponding mimic block 100 to be included in a leakage monitor 24x (or multiple such monitors if more than one is desired). Mimic block 100 receives the same system bias voltage, VDD, as provided to core 14x (see FIG. 1) and from a node 102 that is connected to a source of an n-channel transistor 104. N-channel transistor 104 has its drain connected to an output 1061, of a current mirror 106 and its gate connected to the output of an amplifier 108. The non-inverting input of amplifier 108 is connected to receive the system bias voltage VDD and the inverting input of amplifier 108 is connected in a feedback manner to node 102; thus, one skilled in the art will appreciate that this configuration therefore causes the voltage VDD at the non-inverting input to be duplicated at node 102. Returning to mimic block 100, it also preferably receives the system back bias voltage, VBB, for application to desired back gates therein, and also a clock signal from an output of an OR gate 110. OR gate 110 has two inputs, the system clock signal CLK from bus CB (see FIG. 1) and a STOPCLK signal, which is further described later.
Continuing the description of FIG. 2a, current mirror 106 receives a voltage source signal designated VDDA, which preferably is the source supply for any related analog circuitry and which is also used to supply voltage to amplifier 108. Voltage VDDA is larger than VDD so as to properly achieve the functionality detailed later. Current mirror 106 includes another output 1062 connected to a current mirror 112 and another output 1063 that may be contacted by the switching end of a switch 114. The switching end of switch 114 also may be connected to a node 116 that is also connected to current mirror 112. Switch 114 is further connected to a node 117, where between node 117 and ground is connected a capacitor 118, and where node 117 is further connected to an input 1201 of a comparator 120. A second input 1202 of comparator 120 is connected to receive a reference voltage, Vref. The output of comparator 120 is connected to a controller 122 and, more particularly, to control the starting and resetting of a counter 124 in controller 122. Controller 122 also receives the system clock CLK from clock bus CB, while it provides a switch control signal SW_CTRL to switch 114 and the STOPCLK signal to OR gate 110. Lastly, controller 122 outputs a signal fLx, which is intended to represent either fL1 or fL2 shown in FIG. 1, but is shown more generically as fLx in FIG. 2a so as to correspond to the generic indication of the respective leakage monitor 24x. Accordingly, fLx may be monitored to provide a signal representing the amount of leakage current in monitor 24x, as further appreciated below.
The operation of leakage monitor 24x is now explored in connection with both FIGS. 2a and 2b. Introducing then FIG. 2b, it illustrates a plot of a voltage, VC, across capacitor 118, over time. With transistor 104 operating, a current I1 passes through its source/drain conductivity path. Initially, assume that the STOPCLK signal as input to OR gate 110 is de-asserted low, so that OR gate 110 passes, to mimic block 100, the CLK signal at its other input; at this time, therefore, mimic block 110 mimics the active operation of a corresponding core 14x and uses current I1 as its active current. Further, the current I1 is mirrored, by current mirror 106, at output 1062 in the form of current I2. Also at this time, assume that switch 114 is in its upward position, as shown in FIG. 2a. In response, current I2 is mirrored, by current mirror 106, at output 1063 in the form of current I3.1; moreover, that current I3.1 stores charge across capacitor 118 and, thus, the voltage VC across that capacitor begins to rise as shown following time t0 in FIG. 2b. As VC rises, at a time t1 it reaches and thereafter surpasses Vref; at this time, comparator 120 outputs a first state to controller 122, thereby causing counter 124 to begin to count. In the preferred embodiment, counter 124 counts for a predetermined number of system clock CLK cycles, as shown as CH cycles between times t1 and t2 in FIG. 2b. 
Continuing with the operation of leakage monitor 241, at time t2, that is, after CH cycles of CLK, controller 122 asserts the STOPCLK signal high, resets the count in counter 124, and transitions the switch control signal SW_CTRL so that switch 114 moves to its downward position, that is, opposite of that shown in FIG. 2a. At this point, therefore, the assertion of STOPCLK causes OR gate 110 to present a constant high clock signal to mimic block 100, thereby holding it in one state. Thus, mimic block 100 at this point represents the state during stopped operation, that is, it will mimic the leakage of current through the corresponding core 14x. Thus, the current I1 now used by mimic block 100 is representative of current leakage through core 14x. Moreover, again, current I1 is mirrored, by current mirror 106, in the form of current I2 to current mirror 112. Further, current I2 is mirrored, by current mirror 112, in the form of current I3.2, where the voltage VC across capacitor 118 begins to discharge at a rate reflected in current I3.2. Thus, current I3.2 is proportional to the leakage current through mimic block 100. During the time of discharge of voltage VC following time t2, counter 124 accumulates a new count; once the voltage VC across capacitor 118 reaches and thereafter falls below Vref, which is shown at time t3 in FIG. 2b, then comparator 120 changes its output state, thereby stopping the count in counter 124. The total count in counter 124 at time t3 is referred to herein as M counts, as also shown in FIG. 2b. Thus, by time t3, leakage monitor 24x is informed of the time (CH cycles) for capacitor 18 to charge to a certain voltage during active current usage as well as the time (DCH cycles) for capacitor to discharge from that certain voltage during leakage current usage. Given the preceding, one skilled in the art will appreciate that a ratio may be provided between the N counts and the M counts, with the result providing a relative indication of the active use current relative to the leakage current. This ratio is provided by the signal fLx and, thus, indicates an amount of current leakage in mimic block 100 and, hence, also in core 14x. Thus, the ratio effectively a digital value, converted from the analog provision of mirrored currents, to represent leakage current. Further, while FIG. 2a illustrates one such approach, one skilled in the art may ascertain various alternatives.
Returning to FIG. 1 and applying the observations from FIGS. 2a and 2b to a core 14x, then according to the preferred embodiment each leakage monitor 24x preferably operates in a similar capacity, that is, to provide a signal representative either directly or proportionally an amount of current leakage in the respective core 14x. Particularly, note that mimic block 100 is preferably designed and located in manner to reflect a comparable amount of leakage as the leakage that is then occurring in the remaining functional transistors in the corresponding core 14x. In other words, given the relatively small space in which a core 14x is constructed and the large number of transistors therein as well as the various different possible functions of different ones of those transistors, then at any given time some percentage of the overall number of transistors will be in a non-conducting and non-switching state, and those transistors will leak current, thereby causing a commensurate amount of power consumption. Mimic block 100 is therefore designed and constructed to reflect a proportionate indicator of the amount of this leakage current and, therefore, the signal fLx represents not only the amount of current leakage within a leakage monitor 24x, but it likewise represents the amount of current leakage in the corresponding core 14x. Still further, while separate leakage monitors 241 and 242 are shown in the preferred embodiment, in an alternative preferred embodiment a single such monitor may be provided for the entire SOC block 12, or as still other embodiments more than two such monitors may be included. Lastly, note that the signal fLx may be provided in various forms. In one form, it may have a frequency, where the amount of signal transitions over a period of time may be counted and the magnitude of the count thereby provides a digital representation proportional to the amount of current leakage. In another form, the signal may be a ratio between current leakage and active leakage, as discussed above in connection with FIGS. 2a and 2b. 
Turning now to FIG. 3, it illustrates controller 26 in detail, which preferably is constructed and operates to process the signals, fSx and fLx, from both sets of speed capability indicators 221 and 222 and leakage monitors 241 and 242. For sake of discussion, however, the following discussion first considers controller 26 as considering only one set of fSx and fLx from the speed capability indicators 22x and leakage monitor 24x of a single core 14x, where a later discussion contemplates the processing of those signals from both core 141 and 142. Turning to the details of FIG. 3, frequency signal fSx is connected as an input to a divider 32a that divides the frequency of that signal by a value M, and the result is provided to a counter 32b. The system clock signal is connected, from clock bus CB, as an input to a divider 34a that divides the frequency of that signal by the same value M as used by divider 32a, and the result is provided to a counter 34b. The counts from both of counters 34a and 34b are connected as inputs to a comparator 36. Comparator 36 compares these count values to one another so in effect comparator 36 is comparing a measure of the frequency signal representing the speed capability (i.e., counter 32b) with a comparable measure the frequency of the system clock (i.e., counter 34b), where the comparison is simplified by comparing a scaled version (i.e., scaled by M) of each of these values. Note also that if frequency signal fSx is scaled, as described above, by applying including a frequency divider in speed capability indicator 22x, then an appropriate modification is made, such as by doubling the count in counter 32b if a 50% reduction is made in the output of fSx to correspond to the divider in speed capability indicator 22x. In any event, in response to its comparison, comparator 36 outputs a difference signal, D, indicating the difference between the counts in counters 32b and 34b. Note also that in an alternative embodiment counters 32b and 34b and comparator 36 are replaced with an up-down counter, whereby frequency signal fSx causes an advancement of the up-down counter in one direction (e.g., increment), while the system clock from clock bus CB causes an advancement of the up-down counter in the opposite direction (e.g., decrement). As a result, the ultimate count in such an up-down counter also represents the difference signal, D.
Turning to the response to the difference signal, D, it is connected as an input to a voltage adjust circuit 38, which also receives as an input the signal, fLx, from the corresponding leakage monitor 24x. Voltage adjust circuit 38 may be constructed by one skilled in the art to achieve the functionality described in this document, with such construction being selected from various different combinations of available hardware and/or software, including by way of example a state machine in combination with circuitry (e.g., digital filter) to respond to the various inputs and to provide the described outputs. Also in this regard, voltage adjust circuit 38 is shown to provide two digital voltage indication signals, VDD—D and VBB—D, to power supply 18, where as shown below these signals can cause changes in the power supply voltages VDD and VBB, respectively. Moreover, with respect to the latter digital signal, VBB—D, recall from earlier that the preferred embodiment provides one back bias voltage to the p-channel transistors and another back bias voltage to the n-channel transistors in each core 14x. Consistent with these two voltages, therefore, in the preferred embodiment the digital signal VBB—D includes two different digital signals, one to correspond to the back bias voltage for the p-channel transistors and one to correspond to the back bias voltage for the n-channel transistors. Voltage adjust circuit 38 also receives a LOCK signal from power supply 18. Finally, controller 26 includes an additional divider 40a that also receives the system clock signal from clock bus CB, and divider 40a divides the frequency of that signal by a value N; the result is provided a to a counter 40b, which is operable to provide a clear signal (“CLR”) and a STOP signal to both counters 32b and 34b. 
The preferred operation of controller 26 is now discussed. Counter 40b governs each cycle of operation for controller 26 by counting toward a threshold, where this may be achieved by using a decrementing counter which reaches a threshold of zero or by using a wraparound (or modulo) counter that is considered to reach a threshold each time it returns to zero or some other value. In any event, when counter 40b reaches its threshold, the CLR signal is asserted thereby clearing the values in both counters 32b and 34b, after which both counters being counting in response to their respective inputs. The periodicity of CLR being asserted to periodically clear the counters in this manner is adjustable by the value of N given the then-existing clock frequency provided by clock bus CB. For example, if the system clock is at 250 MHz, and N=10, then counter 40b counts at 25 MHz. With this counting rate, a threshold is associated with counter 40b such that once the threshold is reached, CLR is asserted. At the same time, which therefore also continues after counters 32b and 34b are cleared, speed capability indicator 22x reports a frequency fSx, which recall represents the speed, either directly or scaled downward, at which the relatively small circuit in that indicator transitions (or “oscillates”) at the then-existing voltages for VDD and VBB, and that frequency is scaled downward by M and counted by counter 32b. For example, assume during this time period that given the then-existing VDD and VBB, frequency fSx=500 MHz, and assume M=5, then counter 32b counts at a rate of 100 MHz. During the same time, the system clock signal provided by clock bus CB is scaled by M and that frequency is counted by counter 34b. For example, continuing with the assumption that the system clock is at 250 MHz, then counter 34b counts at a rate of 50 MHz (i.e., 250 MHz/5=50 MHz). At some point, also determined by the count in counter 40b, STOP is asserted and the counts in counters 32b and 34b are compared by comparator 36. In the present example, because the count rate of those counters are 100 MHz and 50 MHz, respectively, then the count in counter 32b will be twice that of the count in counter 34b. Thus, these values are compared by comparator 36 and the resulting difference, D, is provided to voltage adjust circuit 38.
In the preferred embodiment, voltage adjust circuit 38 is operable to analyze the difference, D, either alone or in combination with the leakage signal fLx, and to cause an adjustment in the system voltages VDD and VBB in response to this analysis, where the adjustments are made by issuing counterpart digital voltage indication signals, VDD—D and VBB—D, respectively. In general and as appreciated from the following, these adjustments permit the overall operation of a corresponding core 14x to be adjusted so as to improve the performance efficiency, such as by providing a more optimal amount of energy consumption for a given system clock frequency. To appreciate these aspects in greater detail, the following first provides two examples where the leakage signal fLx is not considered, leaving only the speed capability indicator frequency, fSx, and the system clock frequency, as bases for making adjustments to improve energy efficiency. Thereafter, an additional discussion is provided to further include considerations with respect to the leakage signal fLx. Further, once an adjustment is made, CLR is asserted, thereby clearing both counters 32b and 40b. Thereafter, STOP is de-asserted and those counters again begin counting, again in response to fSx and the system clock on clock bus CB, respectively. Next, the above-described methodology is repeated, that is, STOP is asserted and the difference signal, D, is determined. Accordingly, in the preferred embodiment this entire process may be repeated numerous times (or indefinitely) while system 10 is operating. Consequently, there is an ongoing, periodic, and dynamic evaluation of the difference, D, at different times, with each instance representing a potential adjustment to VDD and VBB in response to that difference. As a result, there is continuous management and optimizing of the energy consumption of system 10.
Looking at a first example of the above-introduced operation, assume at a first time, t1, that fSx, as processed through counter 32b, indicates a speed capability of frequency at 500 MHz, while at the same time assume that the system clock frequency, as processed through counter 34b, indicates a then-existing core operational frequency of 250 MHz; recall also that the difference between these two frequencies, after scaling, is reflected in the difference signal, D, to voltage adjust circuit 38. According to the preferred embodiment, these frequencies and, thus, the scaled difference, D, between them, effectively provide a comparison of the then-operating frequency of core 14x (as caused by the system clock) with the potential capability of what frequency that core could operate at if left to transition freely in response to VDD and VBB, as reflected in fSx; in other words, recall that fSx is provided by a speed capability indicator 22x which is designed to mimic the voltage/frequency characteristics of core 14x if it were operated independent of the system clock. In effect, therefore, for the present example, the 500 MHz indicated by fSx represents that system voltages VDD and VBB provide sufficient power to support a capability of 500 MHz of core operation due to the likeness of the core critical path relative to speed capability indicator 22x, even though the core at time t1 is operating at 250 MHz. From this observation, therefore, it is effectively plausible that a lesser voltage may be used to support then then-required core operating frequency of 250 MHz, that is, the current states of the system voltages VDD and VBB as of time t1 are sufficient to drive 500 MHz of operation, when at that time there is only the need to support 250 MHz of operation. As a result, since voltage adjust circuit 38 is made aware of these factors by way of the difference signal, D, then voltage adjust circuit 38 operates to cause an adjustment in one or both of the system voltages VDD and VBB. Particularly in this regard, voltage adjust circuit 38 provides digital values VDD—D and VBB—D to power supply 18, thereby causing power supply 18 to adjust VDD and VBB, respectively, in proportion to the digital indications. In the present example, therefore, by causing a reduction in VDD and/or an increase in VBB, voltage adjust circuit 38 can continue to monitor the frequency signal fSx and the then-existing frequency of the system clock, as both will continue to be reflected by new determinations of the difference, D. Once the difference, D, is sufficiently low or reaches zero, then it is known that the frequency signal fSx is close to or matches the then-existing frequency of the system clock; at that point, therefore, voltage adjust circuit 38 has provided an improved adjustment to VDD and/or VBB, that is, that is, energy consumption has been reduced versus the state where the difference, D, is a larger or non-zero value, yet the critical path of a core 14x may continue to operate at a same frequency while consuming less energy.
While the preceding has demonstrated considering a single value of fSx from a single speed capability indicator 22x, recall in FIG. 1 that controller 26 is connected to receive multiple values of fSx from respective multiple speed capability indicators 22x. With these connections, various alternatives are contemplated within the present inventive scope. In one approach, controller 26 may operate to cause local voltages for VDD and VBB to be supplied to different local cores, that is, one set of VDD and VBB may be caused to be supplied to core 141 based on its speed capability indicator frequency fS1, while another set of VDD and VBB may be caused to be supplied to core 142 based on its speed capability indicator frequency fS2 However, in some implementations, the use of different values of VDD (and VBB) for a single chip such as SOC block 12 may prove too complex or otherwise undesirable, and thus instead controller 26 may implement some type of optimization as between receiving multiple values of fSx. For example, a conservative approach may be, in the instance of receiving multiple values of fSx, determining the difference, D, based on the value of fSx that is closest to the clock frequency; in this manner, any subsequent adjustment will optimize toward that value, without over adjusting which might occur if the value of fSx that were more distant from the clock frequency were used. Other manners of optimizing any adjustment, based on multiple values of fSx, may be ascertained by one skilled in the art.
Recalling that voltage adjust circuit 38 is also operable to analyze the difference, D, in combination with the leakage signal fLx, two examples in that regard are now explored. As a first example, assume at a time t3 that fSx indicates a speed capability of frequency at 500 MHz, while at the same time assume that the system clock frequency indicates a then-existing core operational frequency of 250 MHz, and again the scaled difference between these two frequencies is reflected in the difference signal, D, to voltage adjust circuit 38. Assume further that the leakage signal, fLx, indicates a relatively small amount of current leakage. With respect to this latter indication, note that a threshold (or more than one threshold) may be established as a basis of comparison to the leakage signal, fLx, such that if fLx is below the threshold, then leakage is considered low, whereas if fLx is above the threshold, then leakage is considered high. Returning then to the example wherein leakage is low, then preferably voltage adjust circuit 38 again provides digital values VDD—D and VBB—D to power supply 18, thereby causing corresponding voltage outputs from power supply 18 of VDD and VBB; however, since leakage is low, the preferred embodiment directs a larger reduction in the magnitude of VDD (by changing VDD—D) as compared to the increase in the magnitude of VBB (by changing VBB—D). Indeed, having detected relatively low leakage, it may be preferable to only reduce VDD while leaving VBB unchanged. The preference of a greater adjustment to VDD versus VBB arises from the observation that by considerably increasing VBB, the threshold voltage of the transistors affected may be such that additional leakage may be possible, which would be undesirable. Thus, the more dominant (or sole) change in this example is made by reducing VDD. In contrast, assume as another example, at a time t4, that the frequencies are the same as were the case at t4 (i.e., system clock of 250 MHz, fSx=500 MHz), but at t4 assume that the leakage signal, fLx, indicates a relatively large amount of current leakage. In this case, then preferably voltage adjust circuit 38 again provides digital values VDD—D and VBB—D to power supply 18, but since leakage is relatively high, the preferred embodiment directs a larger increase in the magnitude of VBB (by changing VBB—D) as compared to the decrease in the magnitude of VDD (by changing VDD—D). Indeed, it may be preferable to only increase VBB while leaving VDD unchanged. The preference of a greater adjustment to VBB versus VDD arises from the observation that by increasing VBB, power consumption is reduced both by reducing current leakage and providing a total overall reduction in power available to drive core 14x. Lastly, recalling that controller 26 may operate in alternative embodiments with respect to either a single core or multiple cores when analyzing speed capability of those cores from the signal fSx, then similarly controller 26 may operate in alternative embodiments with respect to one ore more leakage monitor indications and resulting changes may be made to either global or local values of VDD and VBB.
While the preceding has described preferred embodiments that adjust VDD and/or VBB in response to either or both of fSx and fLx, the present inventive scope contemplates that other system parameters that are dynamically alterable also may be changed so as to affect the operation of a critical path and in response to either or both of fSx and fLx. For example, an additional system parameter is the clock speed provided by system clock 20. In other words, in an alternative embodiment that speed may be made to be adjustable in response to either or both of fSx and fLx. To illustrate this point, consider again the earlier example at a first time, t1, wherein fSx indicates a speed capability of frequency at 500 MHz, while at the same time assume that the system clock frequency indicates a then-existing core operational frequency of 250 MHz. Rather than adjusting VDD, or in combination with adjusting VDD, in an alternative embodiment the system clock frequency may be adjusted according to these indications. For example, in one approach, VDD may be left at its then-current level, while system clock 20 is controlled so that the clock speed is increased toward, or all the way to, a value of 500 MHz. In this manner, the core operational frequency is raised toward that of the perceived capable speed of 500 MHz, the latter of which is provided by fSx. Thus, in this approach, assuming VDD is not changed, then energy consumption is not reduced at time t1, but performance is enhanced by increasing clock speed while leaving VDD and VBB undisturbed. Further, this adjustment may be combined with an adjustment to VDD and/or VBB for further optimization as may be desired. Also in connection with the preceding, it is noted that certain considerations may dictate a preference in some implementations of adjusting one system parameter or parameters while not changing others. For example, in contemporary applications, often an original equipment manufacturer or the like may specify certain parameters that may not be changed, for instance desiring that clock speed be left solely in control of their software. In such a case, dynamically altering the system clock as just described may not be preferable or permissible. In other instances, however, it may prove beneficial, particularly for speed demanding applications. Lastly, note that while the present document has described alterations in the system parameters consisting of VDD, VBB, and clock speed, one skilled in the art may ascertain other system parameters that may be dynamically adjusted in response to fSx and/or fLx with beneficial results and consistent with the present inventive scope.
FIG. 4 illustrates a state diagram of an example of operation of controller 26 of FIG. 1 and with the expectation that the reader is familiar with the examples set forth above so as to simplify the remaining discussion. In general, the state diagram of FIG. 4 assumes a steady state of operation where VDD and VBB are at set levels for SOC block 12, and thereafter a change in the frequency of system clock 20 is desired, such as may be the case under control of software or the operating system, and as may occur due to a task switch or the like. This requested change is represented by a state 50, and the request may be to decrease the system clock speed as shown to the right of state 50 or the request may be to increase the system clock speed as shown to the left of state 50. Each of these alternative paths is discussed below.
Looking to the states following an operating system request to decrease the system clock speed, following state 50 is a state 52, during which system clock 20 is controlled so as to achieve the desired decrease in clock speed. Thereafter, the control moves to state 54, in which the difference, D, provided by controller 26 is evaluated to see if it is sufficiently high so as to warrant an adjustment in the system voltage(s). This determination may be made by comparing the difference, D, to a threshold, and that threshold may be as low as zero. If the difference, D, is insufficient to warrant a voltage adjustment, then the state diagram proceeds to a wait state 56; additionally, power supply 18 issues a LOCK signal to voltage adjust circuit 38, which thereby acts as an acknowledgment that a steady state has again been achieved, given the reduction in clock speed from state 52. The LOCK may be used by voltage adjust circuit 38 to assert its STOP signal to counters 32b and 34b, thereby preventing those counters from counting during the time that STOP is asserted. At some point later, the STOP signal may be de-asserted and CLR asserted, at which time control leaves wait state 56 and returns to step 54, to once again determine and evaluate whether the difference, D, is at a sufficient level to warrant a voltage adjustment. If indeed the difference, D, is at such a level, then the control continues to state 58. State 58 represents the change to one or both of VDD and VBB, consistent with the value of the difference, D, and also possibly in view of any leakage, as provided by the appropriate leakage monitor(s) as detailed above. Once such an adjustment is made, the flow continues to wait state 56 for a period of waiting, after which the difference, D, is again evaluated as the flow returns to step 54. Lastly, note in the preferred embodiment that state 52 preferably represents a step taken under control of software, including the operating system of system 10. However, in the preferred embodiment, steps 54, 56, and 58 are implemented in hardware so as to minimize the software interruption.
Looking to the states following a request to increase the system clock speed, following state 50 is a state 60, during which the request is communicated to controller 26. Thereafter, the flow continues to a state 62, at which time controller 26 maximizes VDD and minimizes VBB; by making these adjustments, then initially the voltage difference between VDD and VBB is maximized so as to provide the largest possible drive current to thereby support a maximum rate of speed for a core 14x. Next, after the voltages of VDD and VBB settle, the flow continues to a state 64, during which system clock 20 is controlled, preferably by the operating system in response to the earlier-introduced change from state 50, so as to achieve the desired increase in clock speed. Thereafter, the control flows to state 66, during which controller 26 operates as detailed earlier. As a result, and as shown in the next state 68, controller 26 determines whether the difference, D, is sufficiently high so as to warrant an adjustment in the system voltage(s). Again, this determination may be made by comparing the difference, D, to a threshold. Thereafter, states 70 and 72 make the appropriate changes, if any, in VDD and VBB and wait for the next successive evaluation of the difference, D, in a comparable manner as described above with respect to states 56 and 58.
Having detailed various aspects, including alternatives with respect to, system 10, attention is now returned to yet additional considerations that may be applied as relating to preferred embodiment implementations of a speed capability indicator 22x (e.g., either 221 or 222). Specifically, FIG. 5 illustrates a block diagram of a general architecture that may be used to form speed capability indicator 22x as a ring oscillator. Speed capability indicator 22x preferably includes three (or some other number) like constructed blocks connected in a serial path and with a feedback from the output of the last block in the path to the input of the first block in the path. Thus, in FIG. 5, three such blocks 22xA, 22xB, and 22xC are shown in serial order, that is, with the output of block 22xA connected to the input of block 22xB, with the output of block 22xB connected to the input of block 22xC, and with the output of block 22xC providing the above-detailed output frequency, fSx, and also being fed back to the input of block 22xA. Each of blocks 22xA, 22xB, and 22xC is shown in general to receive and, hence, to respond to, the two system voltages VDD and VBB. Thus, each block operates in response to those voltages and provides an output, and due to the ring nature of the FIG. 5 configuration, a transition from one block in the ring is provided to a next block in the ring which also transitions and so forth continuing in a ring manner, so as to ultimately produce a steady-state or oscillation frequency that is presented in the output frequency, fSx. Further in this regard, the particular design of the schematic within each of blocks 22xA, 22xB, and 22xC is detailed below, where one skilled in the art should anticipate from the earlier discussion of speed capability indicator 22x as a whole that the schematic is selected so as to model a voltage/frequency response similar to one or more critical paths CPx within a core 14x to which speed capability indicator 22x corresponds.
Before proceeding with a discussion of the detailed implementation within each of blocks 22xA, 22xB, and 22xC, a preliminary discussion is noteworthy with respect to the preferred embodiment manner of identifying a data path and its respective voltage/frequency characteristics that blocks 22xA, 22xB, and 22xC will seek to mimic. The preferred embodiment recognizes that often a circuit, such as a core 14x, may have numerous possible data paths that potentially provide the most restrictive delay through the circuit. In other words, certain data paths through the circuit will be suspected as the most restrictive (i.e., causing the most delay), but indeed out of numerous candidate data paths, it may be cost and/or time prohibitive to consider all of these potential data paths in determining which is the most restrictive. Thus, in the preferred embodiment, and preferably using the same design software and hardware that designs core 14x itself, an appropriate algorithm, and having access to the core network node listings or the like, is operable to identify a reasonable number of data paths where each is a candidate that may represent the most restrictive delay through core 14x, that is, the true critical path through core 14x. Such software and hardware are readily ascertainable by one skilled in the art, and will generally include some type of data storage facility to store the network node listings or other machine-readable circuit description as well as a processor and other supporting circuitry so as to run adequate circuit design, analysis, and simulation software. In any event, once these candidate data paths are identified, the preferred embodiment proceeds to select one as the critical path as further described below.
FIG. 6 illustrates a graph having voltage along its horizontal axis and delay along its vertical axis. Further, within the graph are five plots numbered P1 through P5. These five plots represent voltage/frequency characteristics of five data paths in core 14x that are identified by the preferred embodiment as potentially the most restrictive in core 14x. With this identification and then with a simulation of the voltage/frequency characteristics of each such as shown in FIG. 6, then the most restrictive may be readily selected and thereby deemed to be the critical path CPx. For the sake of simplifying the present example, each plot Px is shown to be linear, where in actual implementation such plots may be curved to various extents. Looking in more detail at each plot Px, each is intended to illustrate the voltage/frequency behavior of the five identified data paths through a core 14x, where that behavior may be based on one or both of the system voltages VDD and VBB. Further in this regard, recall from earlier that in the preferred embodiment, speed capability indicator 22x is constructed and operates in response to system voltages VDD and VBB to provide resultant voltage and frequency characteristics that in general mimic the voltage and frequency characteristics of a critical path CPx in a corresponding core 14x. Accordingly, the voltage of the horizontal axis in FIG. 6 represents either or both of these system voltages. Rather than illustrating the plots in terms of frequency, and for sake of simplifying the present discussion, each plot Px is shown with reference to the delay of each such corresponding data path, where delay as known in the art is merely the reciprocal of frequency. Thus, note that a discussion of a voltage/delays characteristic is therefore understood by one skilled in the art to be equally representative of the voltage/frequency characteristics, where of course one has an opposite slope to the other due to the reciprocal relationship between delay and frequency. Accordingly, for each plot Px in FIG. 6, it may be readily appreciated that at lower voltages, there is generally a greater delay for each path than at higher voltages, where stating the same principle with respect to frequency means that at lower voltages, there is generally a lower frequency for the path than at higher voltages.
Given FIG. 6 and the understanding that the preferred embodiment algorithm identifies a group of candidate data paths and provides an expected voltage/delay plot Px for each such path, then also under the preferred embodiment the data path out of those candidates and having the greatest delay is selected as the probable critical path CPx for core 14x. Thus, in the illustration of FIG. 6, the uppermost one of the plots, namely plot P1, represents this chosen data path as its plot suggests the greatest amount of delay. This choice is made because the greatest delay indicates a greater likelihood that the data path corresponding to plot P1 is the most restrictive in the core 14x to which the path belongs and, hence, it is thus most probable to accurately reflect the true critical path in that core. With this selection, the preferred embodiment, and again preferably using an automated flow algorithm that includes a network node listing or the like of the selected data path, then endeavors to provide design parameters for a schematic to be used as a speed capability indicator 22x circuit that provides similar voltage/delay characteristics as represented by the plot (e.g., P1) for the selected data path. The specific schematic to which these design parameters apply is further appreciated below.
FIG. 7 illustrates a schematic of a preferred implementation for each of blocks 22xA, 22xB, and 22xC of FIG. 5, where the above-discussed design parameters are applied to the schematic to ultimately provide speed capability indicator 22x. For sake of example, only one block 22xA is shown in FIG. 7, with an understanding that preferably the same schematic is duplicated in each of blocks 22xB and 22xC so as to provide a ring oscillator as described above with respect to FIG. 5. Looking then to the schematic of FIG. 7, block 22xA includes an input 22xA—IN that is connected to an input of an inverter 80. The output of inverter 80 is connected to both inputs of a two-input NAND gate 82. The output of NAND gate 82 is connected to a node 84. Node 84 is connected to a first terminal of a capacitive element 86, which has its second terminal connected to ground. Node 84 is also connected to a switched RC (i.e., resistor/capacitor) device 88 which, as further described below, has an RC delay that is adjusted in response to a power supply or supplies. Toward this end, device 88 includes an n-channel transistor 88T having its drain connected to node 84 and its source connected to a first terminal of a capacitor 88C, where the second terminal of capacitor 88C is connected to ground. Also with respect to n-channel transistor 88T, its gate is connected to VDD, and optionally its backgate is connected to VBB. As further discussed below, therefore, the extent to which node 84 is loaded with an RC value from device 88 depends on the voltage provided by VDD, and also by that supplied by VBB if it is also included as shown in FIG. 7.
Continuing with the schematic of block 22xA in FIG. 7, node 84 is also connected to an input of an inverter 92. The output of inverter 92 is connected to both inputs of a two-input NOR gate 92. The output of NOR gate 92 is connected to a node 94. Node 94 is connected to a first terminal of a capacitive element 96, which has its second terminal connected to ground. Node 94 is also connected to a switched RC device 98, which is comparable to switched RC device connected to node 84. Thus, device 98 includes an n-channel transistor 98T having its drain connected to node 94, its source connected to a first terminal of a capacitor 98C that has its second terminal connected to ground, its gate is connected to VDD, and optionally its backgate connected to VBB. Node 94 is also connected to an input of an inverter 98. The output of inverter 98 provides the output, 22xA—OUT, of block 22xA.
Having discussed the specific connections of block 22xA, attention is directed to the considerations that make it a preferable design to form a block that is duplicated to provide a speed capability indicator 22x. In this regard, in the preferred embodiment it is desired to have block 22xA include at least one circuit element of each type that is included in the path that is identified as the critical path based on that path's delay/voltage characteristic as discussed above in connection with FIG. 6. In other words, given a critical path, the preferred embodiment seeks to simulate each building block of that path, with the expectation that the total voltage/delay characteristics of that path should derived from the voltage/delay characteristics of each of its components. Thus, to the extent that the identified probable critical path includes logic functions such as AND and OR, those functions are also included individually in the schematic of FIG. 7. For example, if there are one or more AND gates in the critical path, then the AND function is preferably provided in block 22xA by a single AND function. As another example, if there are one or more OR gates in the critical path, then the OR function is preferably provided in block 22xA by a single OR function. Further in this regard and with respect to the AND function as well as the OR function, generally it is understood that the counterpart NOR function has a maximum delay for a low transitioning signal and the NAND function has a maximum delay for a high transition signal; thus, the NAND and NOR functions, rather than the counterpart OR and AND functions, are implemented in block 22xA, as shown by NAND gate 82 and NOR gate 92 in FIG. 7. With respect to the inversions in the critical path, they are represented in FIG. 7 by inverters 80, 90, and 102.
Continuing with the preferred design considerations for block 22xA of FIG. 7, note that it also includes various devices so as to simulate the effect of the parasitic resistive and capacitive elements of the identified critical path that block 22xA seeks to mimic. In other words, the components of block 22xA that are discussed in the preceding paragraph have explicit schematic counterparts, albeit in a larger quantity, in the identified critical path CPx. The remaining devices in FIG. 7, however, are instead included to mimic the effect of parasitic effects in the corresponding and identified critical path CPx. For example, recall that node 84 is connected to both a capacitive element 86 as well as switched RC device 88. Looking now in more detail to these devices, as a general matter it may be seen that the capacitive aspects of capacitive element 86 as well as switched RC device 88 are generally connected in parallel between node 84 and ground. With respect to switched RC device 88, its operation may be appreciated by considering two instances. In one instance where the difference between VDD and VBB is relatively large, then transistor 88T will present node 84 with a relatively low resistance, thereby causing a larger amount of the capacitance of capacitive element 88C to load node 84 and to add this larger amount of capacitance to the capacitance of the parallel-connected capacitive element 86. Conversely, in another instance when the difference between VDD and VBB is relatively small, then transistor 88T will present node 84 with a relatively large resistance, thereby limiting the amount of capacitance from capacitive element 88C that loads node 84 and thereby adding a lesser amount of capacitance to the parallel-connected capacitive element 86. Note also that RC device 98 operates in a comparable manner with respect to node 94. Lastly, the overall effect of these capacitance contributions will be more readily apparent from the following operational description of block 22xA of FIG. 7.
The operation of block 22xA of FIG. 7 is now described. Looking first to inverters 80, 90, and 98, the serial path between them and the other gates connected in that path ensure an oscillating signal. As a first example, consider the case when a low signal is provided at input 22xA—IN and thus to the input of inverter 80. In response, inverter 80 outputs a high signal, and that high signal is connected to both inputs of NAND gate 82, thereby causing a low output from NAND gate 82. The low output of NAND gate 82 is inverted to a high by inverter 90 and connected to both inputs of NOR gate 92, thereby causing NOR gate 92 to output a low output. The low output of NOR gate 92 is inverted by inverter 100 to a high at output 22xA—OUT; thus, in this first case, the low signal at input 22xA—IN is converted to a high signal at output 22xA—OUT, thereby providing a transition which may be repeated to provide an oscillating signal (i.e., fSx). As a next example, consider the case when a high signal is provided at input 22xA—IN and thus to the input of inverter 80. In response, inverter 80 outputs a low signal, and that low signal is connected to both inputs of NAND gate 82, thereby causing a high output from NAND gate 82. The high output of NAND gate 82 is inverted to a low by inverter 90 and connected to both inputs of NOR gate 92, thereby causing NOR gate 92 to output a high output. The high output of NOR gate 92 is inverted by inverter 100 to a low at output 22xA—OUT; thus, in this second case, the high signal at input 22xA—IN is converted to a low signal at output 22xA—OUT, thereby providing a transition which may be repeated to provide an oscillating signal (i.e., fSx). Also from the above, it may be seen that by replicating block 22xA in each of blocks 22xB and 22xC and given the feedback path from the output of block 22xC to the input of block 22xA, the signal path will continuously transition back and forth, at a frequency dictated by the delay through each block.
Recalling that the preferred embodiment provides a voltage/delay (or voltage/frequency) characteristic in block 22xA (along with blocks 22xB and 22xC) to mimic that of an identified critical path CPx, the delay imposed by block 22xA is now explored in additional detail. First in this regard, with respect to the logic functions of block 22xA, recall from earlier that an automated flow algorithm per the preferred embodiment provides design parameters in view of the selected plot P1, and note now that these parameters are used to determine the preferred transistor parameters used to construct the schematic of FIG. 7. Further in this endeavor, note that the size of each of the transistors used to construct these devices may be adjusted so as to cause the overall voltage/delay characteristics of block 22xA to contribute to speed capability indicator 22x so that the voltage/delay characteristics of the latter approaches that of the identified critical path CPx. Thus, in response to software simulations these transistor sizes may be automatically ascertained so as to approximate in the FIG. 7 schematic the same delays as existing in the critical path CPx. Further, the delay of block 22xA is also affected by switched RC devices 88 and 98. In this regard, recall that generally a large VDD (or difference between VDD and VBB) causes each of devices 88 and 98 to provide a larger capacitive contribution to a respective node 84 and 94. In addition to that consideration, note now that in general, a large VDD (or difference between VDD and VBB) also would cause the remaining logic circuits within block 22xA to operate faster, if they were not loaded with additional circuitry; however, the inclusion of RC switched devices 88 and 98 negates or offsets at least in part this tendency so as to provide a more balanced voltage/delay characteristics of block 22xA so that it better approximates that of the corresponding critical path CPx. Specifically, with a large VDD (or difference between VDD and VBB), the tendency of the logic gates to increase operational frequency is negated in part by the larger capacitive contribution by switched RC devices 88 and 98. Conversely, with a lesser VDD large VDD (or difference between VDD and VBB), the tendency of the logic gates to decrease operational frequency is negated in part by the reduced capacitive contribution by switched RC devices 88 and 98. These offsets, therefore, effectively permit the slope of the voltage/delay characteristics of block 22xA to be adjusted so as to be appropriately sensitive to changes in VDD (and VBB), and to thereby acceptably mimic the same voltage/delay characteristic changes that occur in corresponding critical path CPx in response to changes in VDD (and VBB).
The preceding illustration and discussion of FIGS. 5 through 7 demonstrate various criteria and software simulation for designing a speed capability indicator 22x, but recall further that in the preferred embodiment speed capability indicator 22x is a tunable circuit. More specifically and as is now explored in greater detail, provisions are made so that after the semiconductor device that includes the relevant core(s) and speed capability indicator 22x is constructed, it may be tested and additional tuning (or “trimming”) may be made to the circuit so as to further refine its voltage/delay (or voltage/frequency) characteristics to duplicate those of a corresponding critical path CPx. Note that such post-fabrication testing and trimming may be performed for each individual integrated circuit, or possibly more practical in many instances may be an evaluation of a small percentage of integrated circuits in a larger group or lot, with adjustments made to all circuits in that group based on an analysis of the distribution of differences between the voltage/delay characteristics of the critical paths and speed capability indicators for the evaluated small percentage of integrated circuits. In either event, the preferred manner of post-fabrication trimming is further explored below.
FIG. 8a illustrates a graph having voltage along its horizontal axis and frequency along its vertical axis; thus, in contrast to FIG. 6 described above, FIG. 8a illustrates frequency, rather than its reciprocal of delay. Further, FIG. 8a is intended to illustrate the test-measured voltage/frequency characteristics of two circuits, namely, a first plot CPP is the voltage/frequency characteristic of a critical path CPx and a second plot PF1 is the voltage/frequency characteristic of a speed capability indicator 22x that is attempting to match the characteristics of that critical path CPx. Note that if fabrication occurred in an ideal manner, then plot PF1 in FIG. 8a will correspond to the reciprocal of the voltage/delay of plot P1 from FIG. 6 since plot P1 in FIG. 6 was chosen as the targeted voltage/delay characteristics for speed capability indicator 22x. However, due to various factors known in the semiconductor fabrication art, the actual post-fabrication voltage/delay or voltage/frequency characteristics of speed capability indicator 22x may differ from the intended characteristics.
Continuing with an examination of FIG. 8a, it is further intended to illustrate an example wherein, despite the best efforts of the design criteria and software simulations and algorithms mentioned above, a difference occurs between plots CPP and PF1. Thus, while the intended goal was a match of these two plots, and hence the voltage/frequency characteristics represented by each, the illustration represents a difference between the slopes as well as the magnitude of delay for each plot. This difference is not necessarily drawn to scale, but instead it is shown to an extent to better illustrate the present aspect. In any event, due to the difference, without any further steps per the preferred embodiment, then reliance on the speed capability indicator 22x that operates per plot PF1 to predict the behavior of the critical path CPx that operates per plot CPP would lead to an inaccuracy in that prediction because the two plots differ to a certain extent. According to the preferred embodiment, however, the inclusion of a mechanism for post-fabrication tuning or trimming permits a reduction in this difference, thereby further enhancing the performance of the energy control that occurs in response to the indication of speed capability indicator 22x, as further appreciated below.
In the preferred embodiment, the difference between plots CPP and PF1 is tuned with adjustments corresponding to segments of those two plots, as is now further explored in connection with both FIG. 8a and FIG. 8b. Particularly, in FIG. 8a, four different voltages, V1, V2, V3, and V4 are indicated, where those voltage may be chosen in various manners such as by identifying equally spaced voltage regions along the expected voltage operational range for critical path CPx. Between each adjacent pair of these voltages, each plot has a corresponding segment. For example, between voltages V1 and V2, plot CPP has a segment CPPS1 and plot PF1 has a segment PF1S1. Looking now to FIG. 8b, the same voltages and segments are shown, but FIG. 8b illustrates the segments of plot PF1 after each has been tuned so that each segment much more closely approaches the respective segment of plot CPP. The slope of each segment PF1S1, PF1S2, and PF1S3, of plot PF1 is unchanged, but each such segment is moved closer to the corresponding segment in plot CPP. In the preferred embodiment, this movement of each segment is achieved using a numeric offset for each segment. Specifically, with the plots in FIG. 8a, the frequency difference between two plots at a given voltage is readily obtainable. For example, assume that a frequency difference FD4 exists at voltage V4 in FIG. 8a as between plots CPP and PF1. Further, recall from FIG. 3 that the preferred embodiment measures frequency by counting frequency transitions through counter 32b over a period of time. Thus, in the preferred embodiment the difference FD4 in frequency between the two plot segments at voltage V4 is represented by a count that would occur if the difference in the two frequencies were counted over the same time period as a cycle of counts by counter 32b. In other words, assume at voltage V4 that plot CPP has a frequency f4.1 and assume also at voltage V4 that plot PF1 has a frequency f4.2. Thus, the difference between frequencies f4.1 and f4.2 may be determined, and that frequency if counted over a period of time corresponds to a numeric count. According to the preferred embodiment, this count is used as a numeric offset, such as by adding it to counter 32b by way of example or counter 32b could be pre-loaded with that count each time it is reset. Alternatively, where the preferred embodiment counting is achieved with an up-down counter as mentioned earlier, then the offset may be used as an initial value in the up-down-counter. In any case, therefore, this offset number may be stored into a programmable storage device or the like so that it may included in the total count. Looking at a specific example of an offset that may be implemented in this manner, in the FIG. 8a case illustrated for segment PF1S3, it is seen visually that its depicted frequency is lower than the actual frequency of the corresponding critical path as shown in segment CPPS3; thus, the numeric offset corresponding to the frequency difference determined between the two at voltage V4 is added to the count in counter 32b when the system voltage (or voltage difference) at issue is between voltages V3 and V4. By operating in this manner, segment PF1S3 of FIG. 8a is effectively adjusted and is shown as an adjusted segment PF1S3A at the raised location shown in FIG. 8b. Thus, in FIG. 8b, there is a closer match of segment PF1S3A to segment CPPS3, that is, the voltage/delay characteristics provided by segment PF1S3A are indeed closer to segment CPPS3 in FIG. 8b as compared to FIG. 8a, indicating that the speed capability indicator 22x, as offset and reflected by segment PF1S3A, is very closely predicting the voltage/frequency characteristics of the critical path CPx represented by segment CPPS3.
Completing the discussion of FIGS. 8a and 8b, note that the numeric offset mentioned above is preferably repeated for each set of segments between a same voltage range, preferably by evaluating the offset at the higher voltage for the set of segments. Thus, for segments CPPS2 and PF1S2 between voltages V2 and V3, each segment at voltage V3 has a frequency f3.1 and f3.2, respectively, with a difference between those frequencies. Per the preferred embodiment, this difference is determined as a numeric count that represents that frequency if counted over a period of time, and that count is used as an offset to add to the counts provided by speed capability indicator 22x for voltages between V2 and V3. Thus, segment PF1S2 of FIG. 8a is effectively adjusted and is shown as an adjusted segment PF1S2A at the raised location shown in FIG. 8b, again providing a closer match between the predicted performance as provided by segment PF1S2A and the actual performance of critical path CPx as represented by segment CPPS2. From the preceding, one skilled in the art will appreciate that the same methodology is used for segments PF1S1 and CPPS1 between voltages V1 and V2, and indeed this methodology also may be used for any number of additional segments if a greater number of ranges are desired.
From the above, it may be appreciated that the preferred embodiments provide an integrated circuit speed capability indicator that under operation provides voltage/delay (or voltage/frequency) characteristic that mimic the same characteristics of a corresponding data path, such as a critical path in a larger circuit such as a circuit core. With this indicator, the expected operational attributes of the corresponding data path may be predicted, as may be helpful in various contexts. In one particularly beneficial context, the integrated circuit speed capability indicator is used in an integrated circuit dynamic parameter management structure and methodology that may be used with numerous electronic circuits. Indeed, the preferred embodiments have particular application to improve performance control in circuits with decreased transistor geometries. More particularly, and as detailed earlier, as transistor geometries decrease, it has been observed by the present inventors that a global reduction in VDD as frequency decreases, as implemented in the prior art, may actually increase overall energy consumption, such as by increasing device leakage. In contrast, certain preferred embodiments provide for alterations in device parameters, such as both VDD and VBB, including in view of the then-resulting leakage, so these adjustments are made to counteract any tendency toward increased leakage. As a result, greater improvements in energy consumption can be achieved. As another benefit, greater performance and power efficiency may be realized for a given device size and complexity. As still another benefit, battery-powered devices may support complex operations with a lesser penalty on energy consumption, having application in numerous contemporary devices including those with multiple cores, those support multimedia applications, and so forth. As still another benefit, in alternative embodiments other device parameters may be adjusted in response to a dynamic evaluation of critical path speed capability and/or device leakage, where by way of example one such parameter is an adjustment of system clock speed.
In addition to the preceding benefits, the preferred embodiments also have benefit in devices having varying operational characteristics in a single yield of such devices. In other words, it is known in the manufacturing art that for a group of devices, certain ones of those devices may perform more optimally than others. The prior art uses various approaches to deal with this reality, including by way of example setting certain operational characteristics for the entire group as limited by the lesser performing devices. However, at least with respect to control of various system parameters (e.g., VDD, VBB, clock speed), the preferred embodiments may be implemented in an entire group of such devices, allowing each device in that group to dynamically self-adjust its values of VDD and VBB (or other parameters, including frequency) consistent with the above teachings, thereby permitting different devices from the same group to operate at different system voltages, yielding improved individual operation from device to device as compared to scaling the entire set of devices to accommodate a worst case scenario provided by the lesser performing devices in the yield. Thus, the optimal VDD and VBB may differ from chip architecture to chip architecture, from process node to process node, and also with temperature.
As another benefit of those preferred embodiments that adjust VDD and/or VBB, note that in the past, certain integrated circuit manufacturers have taken a conservative approach in evaluating a lot of devices and then providing operational voltage and frequency specifications according to certain assumptions from the overall results of the lot; often these assumptions typically include worst-case scenarios, such as what performance is achievable from the lower quality silicon in the lot with a fixed voltage supply. As such, often a resulting operational frequency is specified at a corresponding provided fixed system voltage, with both corresponding to a specified amount of power consumption. Thus, devices from the same lot that may have a higher quality silicon are constrained to operate under these lower performance specifications, when in fact some of these devices are capable of operating more efficiently (e.g., higher speeds and/or less energy consumption). However, with the preferred embodiments, and given the ability to dynamically adjust the values of VDD and/or VBB and/or clock speed in response to operational speed capacity and current leakage, then at the same nominal amount of power consumption of the fixed voltage case, greater operational frequencies may be achieved.
From the above, the preferred embodiments provide an overall improvement in not only power management as it relates to provided voltage, but also to energy efficiency in that leakage is also improved and also to performance in that higher frequencies may be achieved at a same level of power consumption used by slower performing devices in the prior art. As another benefit of the preferred embodiments, it has been shown that various substitutions, modifications or alterations could be made to the descriptions set forth above without departing from the inventive scope. Consequently, while the present embodiments have been described in detail, that inventive scope is further intended to include the variations shown herein by example and ascertainable by one skilled in the art, as further is defined by the following claims.

## Claims

1. A single integrated circuit, comprising:
a first circuit having a data path, the first circuit consisting of a first number of logic gates for performing a plurality of logic functions;
a circuit for indicating a potential speed capability of the data path, the circuit for indicating comprising:
a second number of logic gates for providing voltage and delay characteristics comparable to the data path, wherein the second number is less than the first number;
additional circuitry for representing parasitic characteristics in the data path, providing a first capacitive load to the second number of logic gates in response to a first voltage potential applied to the second number of logic gates and to the additional circuitry, and providing a second capacitive load to the second number of logic gates in response to a second voltage potential applied to the second number of logic gates and to the additional circuitry;
wherein the first capacitive load is greater than the second capacitive load when the first voltage potential is greater than the second voltage potential; and

trimming circuitry, adjustable after manufacture of the circuitry for indicating a potential capability, for adjusting the operational speed of the circuitry for indicating a potential speed capability.

2. The single integrated circuit of claim 1 wherein the additional circuitry is for reducing an operational speed of the second number of logic gates in response to an increase in a voltage potential applied to the second number of logic gates and to the additional circuitry.

3. The single integrated circuit of claim 2:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors.

4. The single integrated circuit of claim 2:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage difference for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors and a backgate voltage applied to the selected ones of the plurality of transistors.

5. The single integrated circuit of claim 1:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors.

6. The single integrated circuit of claim 1:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage difference for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors and a backgate voltage applied to the selected ones of the plurality of transistors.

7. The single integrated circuit of claim 1 wherein the trimming circuitry comprises circuitry for representing a frequency adjustment as a digital offset.

8. The single integrated circuit of claim 7:
wherein the trimming circuitry comprises circuitry for representing a plurality of frequency adjustments by a corresponding set of a plurality of digital offsets; and
wherein each digital offset in the plurality of digital offsets corresponds to an operational range of voltage potential to be applied to the second number of logic gates and the additional circuitry.

9. The single integrated circuit of claim 8:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors.

10. The single integrated circuit of claim 8:
wherein the second number of logic gates comprise a plurality of transistors; and
wherein the voltage potential comprises a voltage difference for attaching to at least one of a source node, drain node, or gate node of selected ones of the plurality of transistors and a backgate voltage applied to the selected ones of the plurality of transistors.