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NBS TECHNICAL NOTE 658

U.S. DEPARTMENT OF COMMERCE/National Bureau of Standards

Development of Electric and Magnetic

Near-Field Probes

tx
NATIONAL BUREAU OF STANDARDS

The National Bureau of Standards' was established by an act of Congress March 3. 1901. The Bureau's overall goal is to strengthen and advance (he Nation's science and technology and facilitate their effective application for public benefit. To this end. the Bureau conducts

research and provides: (1) a basis for the Nation's physical measurement system. (2) scientific and technological services for industry and government. (3) a technical basis for equity in trade, and (4) technical services to promote public safety. The Bureau consists of the Institute for Basic Standards, the Institute for Materials Research, the Institute for Applied Technology,

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Environment * — Technical Evaluation and Application * — Fire Technology.

THE INSTITUTE FOR COMPUTER SCIENCES AND TECHNOLOGY conducts research and provides technical services designed to aid Government agencies in improving cost effec- tiveness in the conduct of their programs through the selection, acquisition, and effective

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divisions: Computer Services — Systems and Software — Computer Systems Engineering — Information Technology.

THE OFFICE FOR INFORMATION PROGRAMS promotes optimum dissemination and

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organizational units: Office of Standard Reference Data — Office of Information Activities — Office of Technical

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1 Headquarters and Laboratories at Gaithersburg, Maryland, unless otherwise noted; mailing address Washington, DC. 20234.

3 Part of the Center for Radiation Research.

3 Located at Boulder, Colorado 80302.

« Part of the Center for Building Technology.
I 1 1975

\°$ Development of Electric and Magnetic

f Near-Field Probes

Frank M. Greene

Electromagnetics Division

Institute for Basic Standards

,7t National Bureau of Standards

Boulder, Colorado 80302

y

T2

°*en> of *

U.S. DEPARTMENT OF COMMERCE, Frederick B. Dent, Secretary

NATIONAL BURE.AU OF STANDARDS Richard W Roberts Director

Issued January 1975
Library of Congress Catalog Card Number: 74-600200

National Bureau of Standards Technical Note 658

Nat. Bur. Stand. (U.S.), Tech Note 658, 53pages (Jan. 1975)

CODEN: NBTNAE

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402

(Order by SD Catalog No. 013.46:658) $1.10
CONTENTS

Page

1. INTRODUCTION 2

2. THE ELECTRIC NEAR-FIELD PROBES 4

2.1 General 4

2.2 Electrical Characteristics 6

2.3 The Response of a Short Dipole Antenna 9

2.4 Dipole Antenna Calibration and Measurement Errors 12

3. THE MAGNETIC NEAR-FIELD PROBES 17

3.1 General 17

3.2 Electrical Characteristics 19

3.3 The Response of a Small Loop Antenna 21

3.4 Loop Antenna Calibration and Measurement Errors-- 2 2

4. THE ELECTRIC-DIPOLE RESPONSE OF A LOOP ANTENNA 26

5. PARTIAL -RESONANCE EFFECT IN A LOOP ANTENNA 32

6. ERRORS DUE TO FIELD HARMONICS 36

7. THE NBS CONDUCTING-PLASTIC TRANSMISSION LINE 38

7.1 Introduction 38

7.2 Conducting-Plastic Materials 39

7.3 Construction of the Conducting-Teflon Transmission Line 40

7.4 Electrical Characteristics of the ConductingTeflon Line 41

Conductor Resistance 41

Mutual Capacitance 41

Propagation Function 41

Attenuation Function 42

Phase Function 42

7.5 Transmission Line Attenuation 42

8. CONCLUSIONS 46

9. REFERENCES 47

in
LIST OF FIGURES

Page

Figure 1. Construction details of the 5-cm and 10-cm

dipole antennas 5

Figure 2. Schematic diagram of the portable dipole probe- 7 Figure 3. Equivalent circuit of a short dipole antenna 8

Figure 4. Increase in the effective length and apparent

self -capacitance of a short dipole antenna

with electrical half-length 13

Figure 5. Increase in the effective length of the 2. 5-cm,

5-cm and 10-cm dipole antennas with frequency-- 14

Figure 6. Increase in the apparent self -capacitance of

the 2. 5-cm, 5-cm and 10-cm dipole antennas

with frequency 15

Figure 7. Calibration of the 10-cm dipole probe for

USAFSAM, Brooks AF Base (f = 10 MHz) 16

Figure 8. Construction details of the 3.16-cm and 10-cm

diameter loop antennas 18

Figure 9. Schematic diagram of the portable loop probe 20

Figure 10. Equivalent circuit of a small loop antenna 20

Figure 11. Relative response of the 1-cm, 3.16-cm and

10-cm diameter loop antennas 24

Figure 12. Calibration of the 10-cm diameter loop probe

for USAFSAM, Brooks AF Base (f = 10 MHz) 25

Figure 13. Equivalent lumped circuit of an electrically

small, singly loaded loop antenna showing the

electric- dipole and magnetic-dipole responses-- 26

Figure 14. Worst-case measurement error of a circular-loop

antenna due to its electric-dipole response 30

Figure 15. Worst-case measurement error of the 1-cm,

3.16-cm and 10-cm diameter loop antennas due

to their electric-dipole response 31

IV
Figure 16.

F i gu r e 17.

Figure 18

.

Figure 19 .

Table I.

Table II.

Table III

Table IV.

Table V.

Table VI.

LIST OF FIGURES (Continued)

Page

Simplified equivalent circuit of a small loop

antenna 3 2 Increase in the response of a loop antenna due

to partial resonance 34

Increase in the response of the 1-cm, 3.16-cm

and 10-cm diameter loop antennas due to

partial resonance 35

Attenuation and phase-shift of the conductingplastic transmission line 45

LIST OF TABLES

Page

Dimensions of the 5-cm and 10-cm Dipole

Antennas 4

Approximate Values of the 5-cm and 10-cm

Dipole Capacitances 9

Increase in the Effective Length of a Dipole

with Electrical Length 10

Increase in the Quantity C4/C a of a Thin

Dipole with Electrical Length 11

Dimensions of the 3.16-cm and 10-cm Loop

Antennas 17

Approximate Self -Resonant Frequencies of the

Loop Probes 33

v

DEVELOPMENT OF ELECTRIC AND MAGNETIC NEAR-FIELD PROBES

Frank M. Greene

ABSTRACT

This publicat

design of small el

for measuring haza

and 100 A/m, respe

signed to be used

30 MHz, and consis

small, single-turn

ure the electricrespectively . The

various researcher

radiation -exposure

effects of hazardliving tissue, ele

tile fuels.

ion describes the development and

ectric and magnetic near-field probes

rd-level fields up to 20,000 V/m

ctively. They were originally deover the frequency range from 10 to

t of short dipole antennas and

, balanced, loop antennas to measand magnetic-field components,

probes are intended for use by

s in their electromagnetic,

programs for determining the

level, non-ionizing, EM fields on

ctro-explosive devices, and volaIn order to later extend the use of the probes

to frequencies above 30 MHz, a detailed analysis was

made of several types of measurement errors likely

to be encountered. The principal errors result from

a variation with frequency in: the effective length

and impedance of the dipoles; and the electric-dipole

response and partial resonance of the loops. Corresponding corrections are given for each type of

error as a function of the operating frequency from

10 to 1000 MHz, and as a function of the physical

and electrical sizes of the probes.

As a result of the analysis, the dipoles can now

be used for measurements at frequencies up to 750 MHz,

and the loops to 75 MHz with an estimated uncertainty

of 0.5 dB. Applying recommended corrections will

provide a substantial further increase in the usable

frequency range.

Keywords

Electric near-field probe; electromagneticfield hazard; field-strength measurements; magnetic

near-field probe; near-field measurements; r-f hazard

measurements; semiconducting transmission line.
1. INTRODUCTION

This publication describes the development of portable,

electric and magnetic near-field probes by the National Bureau

of Standards for other U.S. Government Agencies over the past

five years. These probes are designed for use in measuring

hazard- level , electric and magnetic near fields up to 20,000

volts per meter and 100 amperes per meter, respectively, over

the frequency range from 10 to 30 MHz and higher as will be

discussed in detail. The probes are intended for use by

various researchers in their electromagnetic radiationexposure programs for determining the effects of hazard-level,

non- ionizing , EM fields on living tissue, electro -explosive

devices, and volatile fuels.

The probes consist of short, dipole antennas (which measure the component of the electric field parallel to the dipole

axis), and small, single-turn, balanced loop antennas (which

measure the component of the magnetic field parallel to the

loop axis). Each probe has a silicon- junction , semiconductor

diode connected internally across its gap to rectify the induced r-f voltage. The d-c output of each probe is transmitted over a special non-metallic, high-resistance, transmission line (developed at NBS) to a remote d-c electrometer

voltmeter, located up to 30 feet away. Such a resistance line

is essentially "transparent" to the field being measured and

substantially reduces the measurement error that would otherwise result from distortion of the field and r-f pickup if a metallic line were used.

It is known that both the amplitude and phase of an

electromagnetic field have a much more rapid spatial variation

This work was sponsored through interagency agreements with

the USAF School of Aerospace Medicine, Brooks Air Force Base,

Texas 78235, and the National Institute for Occupational Safety

and Health of HEW, Cincinnati, Ohio 45202.
in the near zone (close to the source) than they do in the

far-zone (at a greater distance from the source). In order

to minimize the averaging effect introduced by any finite-sized

probe, it is therefore necessary to make its dimensions electrically small when it is used for near-field measurements.

A probe cannot be used to study a field with a ^structure much

finer than its own dimensions. A small probe also introduces

less perturbation of the field than a larger one, since its

scattering cross-section is smaller.

In order to extend the use of the probes to frequencies

above 30 MHz, a detailed analysis was made of several types

of errors likely to be encountered. The performance of both

the electric dipoles and the loop antennas was investigated

as a function of their operating frequency (from 10 to 1000

MHz) and as a function of their physical and electrical sizes.

Measurement errors, resulting from several causes, have been

analyzed as a function of frequency ,. and corresponding corrections are given in each case to be applied to the measurements.

The errors to be treated result from the following

causes

(a) the variation of the effective- length and internal

impedance of a short, electric dipole as a function

of its electrical length;

(b) the variation with frequency of the electric-dipole

response of a small loop antenna to the electric

field present;

(c) the increase in the response of a loop antenna due

to partial resonance;

(d) the harmonic frequencies present in the fields

being measured;

(e) the use of metallic lines associated with fieldstrength meters.

The practical mks International System of Units (SI) is

used in the principal formulas in this paper.
2. THE ELECTRIC NEAR-FIELD PROBES

2.1 General . The electric-field probes consist of two

electrically-short, dipole antennas 10 -cm and 5-cm in overall

length and are designed to be used for measuring hazard-level,

electric fields in the ranges from 200 to 2000 and from 2000 to

20,000 volts per meter, respectively, over the frequency range

from approximately 10 to 30 MHz and higher. In addition, a third

dipole antenna, having a length of 2. 5-cm, was included in the

theoretical analysis. This size of dipole appears to be

necessary for electric-field measurements at frequencies

above about 750 MHz. It also appears that printed-circuit techniques will probably be required in the design of this dipole

and its associated r-f filter, in order to minimize extraneous

responses

The general construction of the 5-cm and 10-cm dipole

antennas is shown in figure 1 and the dimensions are given

in Table I.

Table I Dimensions of the 5-cm and 10-cm Dipole Antennas

Lengths Gap Width Diameters

L-l L-2 L-3 G D-l D-2 D-3

fern) finches) finches') finches')

5.00 0.5625 0.2500 0.004 0.1250 0.4375 0.3750

10.00 0.7500 0.3750 0.031 0.1875 0.5000 0.3750

Notes: (a) see figure 1 for definitions of symbols;

(b) Teflon dielectric is used in the gap having the

same thickness as indicated above for the gap

width, G.

The dipoles are fabricated in two identical halves as

shown in figure 1. The dipole is formed of two brass rods,

each of which fits into one half of a split, brass center post
G

1— D.

- — -»

3*8}

cs

(a) FRONT VIEW

D

1- L

2

1

G

^6

fVw)

(b) BOTTOM VIEW

Figure 1. Construction details of the 5-cm and 10-cm dipole

antennas. See Table I for dimensions.
and is secured by silver soldering. The inner surfaces of the

gap are then milled to the required dimensions and lapped flat

and parallel to form a built-in gap capacitor. Teflon dielectric is used in the gap, having the same thickness as indicated

in Table I for the gap width, G. A small semiconductor diode

approximately 0.076 inch in diameter by 0.170 inch long is

mounted inside of a polystyrene sleeve, which mounts internally

at the center of the dipole. The diode is connected across

the gap by means of spring contacts formed from the diode

leads . The two halves of the dipole are fastened together

with two 4-40 fillister-head (recessed), nylon machine screws

through the center post as indicated. Solder lugs are provided, as shown, for connecting a filter network.

Each dipole is mounted at the end of a 36 -inch-long

,

tubular, fiber-glass handle approximately 1/2-inch outside

diameter having a 1/16-inch wall thickness. The 3/8-inchdiameter end of the dipole center post slides into one end of

the handle and is secured with two 2-56 flat-head, brass

machine screws. Two twisted pieces of the conducting-tef Ion

monofilament, described in Section 7, connect the balanced,

dipole- filter output at the probe end of the tubular handle

to a military- type PJ-291 miniature, twin plug secured

internally at the other end.

2.2 Electrical Characteristics . A schematic diagram of the

portable dipole probe is shown in figure 2. Each probe originally had a type 1N4148 silicon- junction , semiconductor diode

connected internally across its gap, as shown, to rectify the

induced r-f voltage. Type 1N5711 Schottky-barrier diodes

were used in the later models because of their improved

response at frequencies to 1000 MHz. A balanced r-c, r-f
IDO

o

Q

00

R, = R 2 = 100 kA

C, S8.5 or 32 pF (BUILT-IN)

C 2 = 500 pF

ELECTROMETER

VOLTMETER

Figure 2. Schematic diagram of the portable dipole probe.

filter, consisting of R-l, R-2, C-l, and C-2, is built into

the tubular handle next to the probe to prevent any r-f voltage, picked up on the resistance line, from being rectified

by the diode and causing an error in indication. Capacitor

C-l is built into the gap at the center of the dipole, as

mentioned in Section 2.1. The resulting value of capacitance

is determined by the width of the gap and the type of dielectric material used. The design value selected for C-l largely

determines the sensitivity of the probe as will be discussed

further in Section 2.3.

A 30-foot length of the non-metallic, high-resistance

line, described in Section 7, is used to connect the probe

with a remote, high-resistance, d-c electrometer voltmeter.

This meter has an output reading roughly proportional to

electric-field strength over a voltage range from approximately 1 to 10 volts. The line has a resistance of approximately 40,000 ohms per foot, giving a total loop resistance

of 1.2 megohms. The electrometer is used with an input-resisO Q

tance setting of 10 or 10 ohms, so that the error due to

the i-r drop on the line is less than 1 percent. The d-c

voltage level used is sufficient to override any noise present

on the line, as discussed in Section 7.
The detector time constant is such that the probes measure essentially the instantaneous peak value of a steady-state

cw field over the measuring range involved here. This has

been multiplied by 0.707 to convert the indication to rms

(root-mean-square) field strength. For the case of pulsed

2 fields having a duty cycle of 0.05 or greater, the difference

between the cw and the pulse indication is small. The probes

indicate the rms value of the field during the pulse, i.e.,

0.707 times the instantaneous peak value of the pulse.

The equivalent circuit of a short, balanced dipole

antenna is shown in figure 3. The self -impedance of a short

Figure 3. Equivalent circuit of a short dipole antenna.

dipole is essentially capacitive as represented by C . The

a gap capacitance across the center terminals is represented by

C, , which includes the diode capacitance. V. is the r-f voltage induced in the dipole by the electric field, E. V is

the output r-f voltage appearing across the gap and diode at

the center of the dipole as shown. The radiation and ohmic

resistances are considered negligible. The approximate values

of C and C\ for the 5-cm and 10-cm dipoles are shown in

a b

r Table II.

'The duty cycle of a periodically recurring pulse is the ratio

of the pulse width to the period of the pulse.
Table II

Approximate Values of the 5-cm and 10-cm Dipole Capacitances

Dipole

Length

fern)

Capacitance

C C, a b

fpFl (PF)

5.00

10.00

0.5 32

0.7 8.5

2.3 The Response of a Short Dipole Antenna . The voltage, V.,

induced in a short electric dipole by the component of the

electric field, E, parallel to the axis of the dipole is

given by

,V

i - E • L e£f (1)

where L rr is the effective length of the dipole, in meters,

and is as defined by equation (1) for the receiving case [1]

.

The effective length is a function of frequency and increases with the electrical length of the dipole. For a thin

dipole having essentially a sinusoidal current distribution,

the effective length is given approximately by the following

asymptotic expression [2,3], valid for <_$l < tx:

'eff

tan

2

, meters, (2)

where L = 21 is the overall physical length of the dipole in

meters; 3 = 2tt/A, where X is the wavelength in meters. For an

electrically-short dipole having an overall length, L < 0.1X,

'eff (1/2)L. For a half -wavelength, self -resonant dipole,

L rr = (2/tt)L. Several numerical values derived from equation

(2) are shown in Table III.

'Figures in brackets indicate literature references at the

end of this paper.
Table III

Increase in the Effective Length of a Dipole

with Electrical Length

i/X S£

(radians')

.01

.02

.05

.10

.15

. 20

.25

06283

12566

31416

62832

94248

25664

57080

(degrees ) 3 60

7 20

18 .00

36 .00

54 .00

72 .00

90 .00

0003

0013

0083

0343

0812

1563

2732

The angle 0, in Table III, is the equivalent electrical

half-length of the dipole expressed in degrees (6 = 360 £/A)

The induced voltage, V- , given by equation (1), varies with

the electrical length of the dipole in accordance with equation (2). In the case of the NBS dipoles, the output voltage,

V , is also determined by the capacitive voltage-divider ratio

comprised of C and C, and is given by

V

o

" C + C, a b

Vl (3)

For lossless capacitors, the divider ratio in equation (3) is

independent of frequency, giving V the same frequency depend

ence as V-, so long as C and C, remain fixed in value. The

i ' & a b

built-in capacitive voltage divider and the dipole effective

length are both made use of, in the NBS dipoles, in fixing

their measurement sensitivity. However, the apparent dipole

capacitance, C', increases with frequency, as will be shown,

which will give V a further frequency dependence, depending

on the value of C, . b

10
By making use of Schelkunof f ' s analogy with conventional

transmission-line theory [4], it can be readily shown that the

dependence of the apparent self -capacitance , C', of a thin

dipole on its electrical length, will be given approximately

by the following asymptotic expression, valid for < $& < tt/2

C = C a a tan g£

3&

(4)

where C is the low-frequency self -capacitance of the dipole

ci The increase in the quantity C'/C with electrical length is a a shown in Table IV.

Table IV

Increase in The Quantity C'/C of a Thin Dipole

a a with Electrical Length

i/X 3& e c/co fradians) fdegrees) & <x

.01 .06283 3.60 1.0013

.02 .12566 7.20 1.0053

.05 .31416 18.00 1.0343

.10 .62832 36.00 1.1563

.15 .94248 54.00 1.4604

.20 1.25664 72.00 2.4491

.25 1. 57080 90.00 00

It can be seen from Table IV that as i/X + 0, C' -» C o

. a a For i/X < 0.03, the increase in C over C will not exceed 1 — a a percent. However, when i/X = 0.15, the value of apparent

self -capacitance, C, will be 46 percent larger than the lowa frequency value. When the dipole is one half -wavelength long

{i/X = 0.25), its apparent self -capacitance is infinite, due

to resonance with its self - inductance . The radiation resistance of the order of 72 ohms, in this case, is in series with

these reactances.

11
2.4 Dipole Antenna Calibration and Measurement Errors . The

5-cm and 10-cm NBS dipoles are usually calibrated at a relatively low frequency (f = 10 MHz), such that C = C , and

3- 3. L ££ = L/2. The gap capacity, C, , is large compared to C . When these dipoles are used at higher frequencies, the change

in C' and L ££ with frequency will represent a measurement

error. The percent error in the measurement will be approximately equal to the sum of the percentage increases in C and

L rr from their 10 MHz calibration values. This can be deter- e±±

mined from equations (2) and (4), or Tables III and IV (for

the case when C, >> C )

.

b a

J The worst-case measurement errors for a short dipole,

due to the above causes, are shown plotted vs. the electrical

half-length, l/X, in figure 4, and vs. frequency in figures 5 and 6 for the 2. 5-cm, 5-cm, and 10-cm dipoles. As can be seen,

the total measurement error will be approximately 0.5 dB (6 percent) when the overall length of the dipole is 1/8 wavelength

(Ji = A/16) . This would correspond to a frequency of 375 MHz

for the 10-cm dipole, 750 MHz for the 5-cm dipole, and 1500 MHz

for the 2. 5-cm dipole. When it is practicable to calibrate the

dipoles at each frequency used, these errors can be greatly

reduced.

Typical calibration curves, made at a frequency of 10 MHz,

for the 10-cm dipole probe are shown in figure 7 at the end of

this section. The same curves can be used for the 5-cm dipole,

except that the values of electric-field strength obtained

from the curves should be multiplied by a factor of 10. The

calibrations were made at NBS in terms of a high-level standard

electric field, generated in a special parallel-plate stripline. This was supplemented by the use of the standard

(receiving) antenna method of calibration [5]

.

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W)O

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w/A *H19N3dlS ai3ld-Dld!3313 SWd (0)

16
3. THE MAGNETIC NEAR- FIELD PROBES

3.1 General . The magnetic-field probes consist of two

electrically-small, single-turn, balanced loop antennas 10 -cm

and 3.16-cm in diameter, and are designed to be used for measuring hazard-level, magnetic fields. The response of the loop

antennas is directly proportional to frequency, but averages

from roughly 0.5 to 5.0 and from 5.0 to 50 amperes per meter,

respectively, over the frequency range from approximately 10

to 30 MHz or higher.

In addition, a third loop antenna, having a diameter of

1-cm, was included in the theoretical analysis. This size of

loop appears to be necessary for magnetic-field measurements

at frequencies above about 75 MHz. It also appears that

printed-circuit techniques will probably be required in the

design of this loop and its associated r-f filter, in order

to minimize extraneous responses.

The general construction of the 3.16-cm and 10 -cm loop

antennas is shown in figure 8, and the dimensions are given

in Table V.

Table V

Dimensions of the 3.16-cm and 10-cm Loop Antennas

Diameters Gap Widths Lengths

D-l D-2 D-3 D-4 G-l G-2 L-l L-2

(cm) (inches) (inches) (inches)

3.16 0.1250 0.4375 0.375 0.001 0.031 0.5625 0.250

10.00 0.1875 0.6250 0.375 0.001 0.031 0.8750 0.375

Notes: (a) see figure 8 for definitions of symbols;

(b) Mylar dielectric 0.001" thick is used in gap G-l

at the base of each loop.

17
(a) FRONT VIEW

(b) BOTTOM VIEW

Figure 8. Construction details of the 3.16-cm and 10-cm

diameter loop antennas. See Table V for dimensions.
The loop antennas are fabricated in two identical halves

as shown in figure 8. The loop is formed of two semicircular

brass rings, each of which fits into one half of a split,

brass center post and is secured by silver soldering. The

inner surfaces of the gap are then milled to the required

dimensions and lapped flat and parallel to form a built-in

gap capacitor. Mylar dielectric film 0.001 inch thick is

used in the gap (G-l) , as indicated in Table V. A second

insulated gap 0.031 inch wide is provided at the top of each

loop as shown. A small semiconductor diode is mounted inside

of a flanged polystyrene sleeve, which mounts internally in

this gap in a similar manner to that previously described for

the dipoles in Section 2.1. The two halves of the loop are

fastened together with two 4-40 fillister-head (recessed),

nylon machine screws through the centerpost as indicated.

Solder lugs are provided, as shown, for connecting a filter

network.

Each loop is mounted at the end of a 36-inch-long tubular,

fiberglass handle. The loops can be either rigidly fastened

to the handle or made to swivel to facilitate aligning the

loop axis with the field vector when making measurements

.

Conducting-teflon monofilaments connect the probe output at

one end of the handle to the twin plug at the other, as previously described for the dipoles.

3.2 Electrical Characteristics . A schematic diagram of the

portable loop probe is shown in figure 9. Each probe originally had a type 1N4148 silicon- junction, semiconductor diode

connected internally across its gap, as shown, to rectify the

induced r-f voltage. Type 1N5711 Schottky-barrier diodes were

used in the later models because of their improved response at

19
r, = R2 = 100 kA

Ci = C 2 = 500 pF

ELECTROMETER

VOLTMETER

Figure 9. Schematic diagram of the portable loop probe.

the higher frequencies. A balanced r-c, r-f filter, consisting of R-l, R-2, C-l, and C-2, is built into the tubular

handle next to the probe to prevent any r-f voltage, picked

up on the resistance line, from being rectified by the diode

and causing an error in indication. The capacitor, C-l, is

built into the base of the loop, as previously discussed, and

serves as an r-f by-pass capacitor to complete the loop r-f

circuit without short-circuiting the rectified d-c output.

A 30-foot length of resistance line connects the probe with

the remote, high-resistance, d-c electrometer voltmeter, as

in the case of the dipoles.

The equivalent circuit of the small balanced loop antenna

is shown in figure 10. L is the internal, low-frequency

ci

#- +

DC

Output

Vj/2

Figure 10. Equivalent circuit of a small loop antenna

20
inductance of the loop. C represents the combined distributed

capacitance of the loop and the gap and diode capacitances,

and is what determines the partial resonance effect to be discussed in Section 5. The radiation and ohmic resistances are

negligible for small loops in this frequency range and are

being ignored in this treatment. V- is the r-f voltage induced in the loop by the magnetic field, H. V is the output

r-f voltage appearing across the gap and diode at the center

of the loop. At low-frequencies where C can be neglected,

a V

o

s v

3.3 The Response of a Small Loop Antenna . The induced voltage, V-, can be determined from one of Maxwell's equations

relating E and H at any point. For the sinusoidally-timevarying, steady-state case, this is [6,7]

V x E = -jo)uH. (5)

If both sides of equation (5) are integrated over the circular

area of the loop, A = irr 2

, and Stokes' theorem is applied, we

obtain

2tt

f> E-dJl = -jwuHA. (6)

o

£ is the circumference of the loop, I = 2irr, meters. From the

Maxwell -Faraday law, the left-hand side of equation (6) is the

induced voltage, V-, in the loop. So that

2tt $ E«dT = V- = -jcoyHA, volts, (7)

o where w = 2irf, y = 4tt10 , and H is the normal component of

the magnetic-field strength in amperes per meter, assumed constant over the area of the loop. It should be noted from

equation (7) that the induced voltage, V^, lags the magnetic

21
field, H, by 90 degrees. This will be made use of later in

evaluating errors due to the electric-dipole response of the

loop

.

From equation (7) , the magnitude of the induced voltage

can be written

|Vi

| = 0.2Tr 3 fMHz d

2 H, volts, (8)

where fj». is the operating frequency in megahertz, and d is

the mean diameter of the loop in meters. As can be seen, the

response of the loop (at low-frequencies) is proportional to

frequency which somewhat complicates its calibration and use.

The response is also proportional to the area of the loop,

A = 7rd 2 /4, in square meters. The relative responses of the

1-cm, 3.16-cm, and 10-cm diameter loop antennas are shown vs.

frequency in figure 11.

3.4 Loop Antenna Calibration and Measurement Errors . Measurement errors for a small loop antenna due to its electricdipole response and partial resonance effects are treated in

detail in Sections 4 and 5, respectively. Typical values will

be given here for the three sizes of loops under discussion,

as determined from figures 15 and 18. The total worst-case

error resulting from the above effects is approximately 0.5 dB

(6 percent) when the loop diameter is 0.01 wavelength. This

would correspond to a frequency of 25 MHz for the 10-cm diameter

loop, 75 MHz for the 3.16-cm diameter loop, and 250 MHz for

the 1-cm diameter loop. It is possible to greatly reduce these

errors by following an experimental procedure outlined near the

end of Section 4.

Typical calibration curves, made at a frequency of 10 MHz,

for the 10-cm loop probe are shown in figure 12 at the end of

this section. The same curves can be used for the 3.16-cm

22
diameter loop, except that the values of magnetic-field strength

obtained from the curves should be multiplied by a factor of

10. These curves are valid for use at a frequency of 10 MHz

only. When measurements are made at any other frequency, the

value of H obtained from the curves should be. multiplied by

the factor 10/f, where f is the other frequency in megahertz.

The calibrations were made at NBS in terms of a high-level

•standard magnetic field, in a manner somewhat similar to that

described for the electric dipoles in Section 2.4.

23
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24
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25
4. THE ELECTRIC-DIPOLE RESPONSE OF A LOOP ANTENNA

This section will treat the electric-dip ole response of

a loop antenna when measuring H. As is well known, this

response will contribute to the H-field measurement error.

It can be minimized in any of three ways: (a) by the use of

a doubly loaded loop as proposed by Whiteside [8] ; (b) by

making the linear dimensions of the loop small compared to the

wavelength, if a singly loaded loop is used; and (c) by following a procedure, when measuring H, that will average out

the electric-dipole response, as will be discussed later.

The equivalent lumped circuit of an electrically small,

singly loaded loop antenna showing the electric-dipole and

magnetic-dipole responses is given in figure 13. Since the

loop is assumed to be small, the radiation resistance has

been ignored.

Figure 13. Equivalent lumped circuit of an electrically

small, singly loaded loop antenna showing the electricdipole and magnetic-dipole responses.

E

is the induced voltage in the loop due to the electricdipole response to the component of the electric field,

E, in the plane of the loop. VV. = L f-p'E, where L rr is the electrical effective length,

with E.

VF is in time phase

26
Vu is the induced voltage due to the magnetic-dipole response

to the component of H normal to the plane of the loop.

Vo = -jiouH»A. Vu is in time-phase quadrature with H.

Z, = -j/coC, where C is the low-frequency self -capacitance of

the loop. A is the area of the loop. A = 7rd 2 /4.

Z

2 = jooL, where L is the low-frequency self -inductance of the

loop.

Z

T is the load impedance connected to the loop terminals,

i.e., the diode load impedance, the gap capacitance, etc.

The total voltage, V., induced in the loop is the complex

sum of Vu and V

p

, i.e.,

V

E

V

I

The error in measuring H, caused by the electric-dipole response, can therefore be determined from the ratio of the

voltages produced across the load by VY and VY,. Call these

voltages Vpj and VHT respectively. The error ratio, which

is in general complex, is then given by

(10)

This ratio is independent of the load impedance, Z

T

, as can

be seen from equation (10). The induced-voltage ratio, VE

/Vtr,

is given by

(11)

V

H

•ooyw \ H /

where the electrical effective length, L rr, has been assumed to be equal to three-quarters of the length of one side,

w, for a square loop [9] . Before equation (11) can be evaluted, both the magnitude and phase of the complex wave - impedance

,

E/H, must be known.

27
For the case of a uniform plane wave, where E = 120ttH,

equation (11) becomes

\43w/ — =

where 3 = 2tt/A, and A = free-space wavelength, meters. When

equation (12) is substituted in equation (10) along with

Whiteside's expressions for L and C, his formula for the error

ratio of a small, singly loaded square loop is obtained

(wV/ fl - 4.32 \

X )\Q - 3.17/ !^,3MH. —

Iwhere 0, = 2 log (4w/a) , and a is the radius of the conductor

forming the loop.

Since the bracketed term involving Q, at the extreme right

in equation (13) is approximately unity, if tt is large, the

error ratio for a square loop can be approximated by

^=~ V -j **(*)• d4)

HL \A/

For a circular loop of diameter, d, the error ratio is [10]

-3 V

-HL "(0

where A is the wavelength in meters. As can be seen, the

error ratio is directly proportional to the electrical size

of the loop, w/A or d/ A . It can be seen from equations (14)

and (15) that the error ratio for a circular loop is approximately 3.5 dB less than for a square loop, when d = w.

For the plane -wave case in which E and H are in time

phase, the error ratio given by equation (15) is imaginary,

as shown. In the near- field case in which E and H may

approach a time -phase -quadrature relationship (in the limit

28
as the source is approached) , equation (15) becomes real in

the limit, and also must be corrected for the new ratio E/H

existing in the near zone. In the former plane-wave case, an

error ratio of 0.1 will result in an error due to E-field

pickup of only 0.5 percent. In the near-zone case the resulting error may be as high as 10 percent or more in the limit

for the same error ratio.

If the magnitude and phase angle of the wave impedance

are not known, the following experimental technique can be

used to minimize the error due to E-field pickup. While

making an H-field measurement, the loop is slowly rotated in

its own plane, about its axis. If no E-field pickup is present,

the output indication of the loop should not change. If a change is noted, the correct value of H can be determined by

taking the average of the high and the low readings as the

loop is slowly rotated through 360 degrees

.

Equation (15) can be used to estimate the worst-case

error in the H-field measurements and is shown plotted vs.

d/X in figure 14. For a value of d/A = 0.1, for example,

this error may be as high as 60 percent. This error is also

shown plotted vs. frequency for the 1-cm, 3.16-cm, and 10-cm

diameter loops in figure 15.

29
Phoo

t-H

>H

rt iH3O

?H . •H o u

on 03 X m <u z o m

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31
5. PARTIAL -RESONANCE EFFECT IN A LOOP ANTENNA

Partial resonance is the result of the combined effect of

the distributed capacitance of the loop and the gap and diode

capacitances, represented by C in figure 10 (Section 3.2).

Figure 16. Simplified equivalent circuit of a small loop

antenna.

Using the simplified circuit diagram shown in figure 16,

the response can be written [11]

V

V.

"J coC J ir

"*vj(-L-» h

+ i^- -^)

(16)

X

where Q = ^r-, X = 2irf L = x R ' o p 2iTf Co o'

and f is the

o self -resonant frequency of the loop. The resistance, R, is

assumed to remain constant with frequency, but turns out to

be negligible anyway, as will be seen.

The magnitude of equation (16) can be rearranged to give

V

V

1 +

2-x 2 0/(1-6')

(17)

where 6 = w/oo is the ratio of the operating frequency to

the resonant frequency of the loop. It can be shown that

32
if Q _> 10, and 6 <_ 0.75, the second term within the bracket

will be negligible, and equation (17) will simplify to

(18)

o V.

i

1 - 6

Equation (18) can be used to estimate the required correction

in the loop measurement for values of 6 up to 0.75, and is

shown plotted in figure 17. The correction is also shown

plotted vs. frequency for the 1-cm, 3.16-cm, and 10-cm

diameter loops in figure 18. Under the above conditions

these corrections are essentially independent of the Q of

the loop (that is, independent of its losses) and depend

only on the ratio, co/co , defined above. The approximate

self -resonant frequencies for the three loop probes are

given in Table VI.

Table VI

Approximate Self -Resonant Frequencies of the Loop Probes

Loop Diameter Frequency

10.00 cm

3.16 cm

1.00 cm

280 MHz (a)

760 MHz (b)

2060 MHz (c)

Note : (a) and (b) are measured frequencies. (c) is

projected from (a) and (b)

.

33
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34
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35
6. ERRORS DUE TO FIELD HARMONICS

If harmonics are present in the fields being measured,

they can contribute to the resulting measurement error. The

effect of field harmonics in the case of electric-field measurements made with short, dipole antennas is not as serious

as it is in the case of magnetic-field measurements made

with small, loop antennas. This results from the fact that

the principal response of a short dipole does not increase as

rapidly with frequency as does the response of a loop, as has

been shown. The worst-case correction for dipole measurements

can be estimated by adding linearly the magnitudes of the

various field harmonics present, since the detector circuit

is essentially peak reading as mentioned in Section 2.2. The

magnitude of each harmonic should be multiplied by the appropriate factor as determined from figures 5 and 6 in Section 2,

in order to take into account the changing dipole response

with frequency.

As an example, consider a third harmonic field at 120 MHz,

the level of which is -30 dB with respect to a 40 MHz fundamental-frequency field being measured. The magnitude of this

harmonic is roughly 3 percent of the fundamental. In the case

of the dipole antenna, the increase in response at 120 MHz is

approximately 0.6 percent for the 10 -cm dipole and 0.1 percent

for the 5-cm dipole, as can be estimated by extrapolating

figures 5 and 6. Assuming that the harmonic signal adds

linearly to the fundamental, the resulting worst-case measurement error is 3.6 percent for the 10-cm dipole, and 3.1 percent

for the 5-cm dipole due to this cause.

The principal response of a loop antenna is proportional

to frequency, as was shown in figure 11 for the 1-cm, 3.16-cm,

and 10-cm diameter loops, and as was discussed in Section 3.3.

In addition, both the electric-dipole response of the loop,

36
and the partial-resonance effect, previously discussed in

Sections 4 and 5, respectively, also increase with frequency,

which can further accentuate the harmonic error. The harmonics

of many transmitters are at least 30 decibels below the

fundamental-frequency output of the transmitter. If the harmonics are stronger than this, it is possible to use filters

in the transmitter output to further attenuate them, although

many commercial transmitters probably do not take advantage

of this.

If the levels of the harmonic fields are known, the

resulting contribution to the worst-case measurement error,

due to the above three factors, can be estimated by reference

to figures 11, 15, and 18. Since the phase relationships between these three factors is, in general, not known, the worst

that can happen is for them to add linearly to the principal

response of the loop probe at the fundamental frequency.

The apparent magnitude of the harmonic, in the above

example, will be enhanced by a factor of three (from 3 percent

to 9 percent) due to the loop's principal response. The

electric-dipole response and partial resonance of the loop

will further increase this to a worst case error due to this

one harmonic of 14 percent for the 10 -cm loop and 10 percent

for the 3.16-cm loop as can be determined from figures 15 and

18. If the effect of only one additional harmonic were taken

into account, the worst-case errors could be doubled. This,

of course, would be in addition to the other errors discussed.

There is no simple way of accurately evaluating these harmonic

errors unless the actual harmonic content of the field is

known.

37
7. THE NBS CONDUCTING-PLASTIC TRANSMISSION LINE

7.1 Introduction . In the past, it has been found difficult

to accurately measure the absolute strength of electromagnetic

fields having a complex spatial distribution over a significant

range of magnitudes. Part of the difficulty has been in the

use of a metallic r-f transmission line between the dipole

measuring antenna and the receiver, or selective voltmeter,

located at a point remote from the antenna. Such metallic

lines often cause large measurement errors because the line

not only perturbs the field being measured, but the unwanted

r-f currents induced on the line contribute to the total

response of the field-strength meter.

One method of avoiding this difficulty is to use some

form of radio, optical., or acoustical telemetering transmitter

in place of the metallic line. This transmitter, located at

the measuring antenna, can be used to transmit the desired

information to a remote readout unit. This technique has

been employed in various ways by a number of experimenters in

the field. However, because of the nature and complexity of

the telemetering instrumentation, the resulting devices have

had varying degrees of success, but in general have been

seriously limited in the usable range of frequencies and magnitudes over which absolute measurements could actually be

made . The NBS near-zone field-strength meters being described

in this paper avoid these telemetering difficulties by making

use of a special technique. The complex telemetering instrumentation, mentioned above, is eliminated by transmitting the

detected output of the measuring antenna over a completely

non-metallic electrical transmission line which does not

appreciably interact with the field being measured. If the

transmission line is made of sufficiently high-resistance

38
material, it will be essentially "transparent" to the surrounding field. Thus, it will not interact appreciably with the

field being measured or have unwanted coupling with the measuring antenna. This is to say that the r-f currents induced

in the transmission line by the field will be negligibly small,

resulting in insignificant reradiation, or that any r-f energy

propagating along the line will be heavily attenuated because

of the extremely high loss of the line. However, the detected

d.-c output voltage of the measuring antenna can be transmitted

without appreciable loss if a high-resistance d-c vacuumtube voltmeter or electrometer is used as the readout device

at the receiving end of the line [12].

7.2 Conducting-Pla s tic Mat e rials . The high resistance, nonmetallic transmission lines used by NBS employ conductingplastic monofilaments in place of the usual copper conductors.

In the type of material used, the plastic matrix is made conducting to the desired extent by filling (up to 30 percent)

with finely powdered carbon (carbon black) during initial

stages of manufacture while the plastic is in a more or less

liquid or molten form. The finely-divided carbon is dispersed

throughout the plastic to make as homogeneous a mixture as possible. Electrical conduction results from actual physical

contact between the carbon particles suspended in the plastic

matrix. Not all plastics lend themselves to this application.

Only certain types of plastic molecules will properly accept

carbon particles in a manner which renders the material electrically conducting, without a deleterious effect on the final

mechanical properties. For example, nylon is one of the types

which, so far, has been found to be unsuitable by various

manufacturers. Several types of conduct ing-plastic were tested

at NBS for this purpose. The most suitable was found to be

conducting-Teflon made to NBS specifications.

39
7 . 3 Construction of the Conduct ing -Teflon Transmission Line . The parallel -conductor balanced transmission line was constructed using two 0.03 inch diameter nylon-coated, conductingTeflon monofilaments, each approximately 30 feet in length.

Each filament was in turn pulled through a separate 1/8 inch

diameter, vinyl -coated, woven-fiber-glass sleeve for added

strength and insulation. These two 30 -foot lengths were then

pulled together through an equal length of heat -shrinkable

irradiated-polyvinyl chloride tubing, which served as the

outer jacket of the parallel -conductor line. Conducting

silver cement was used to electrically bond the ends of the

conducting-Tef Ion filaments to the special twin-connector

plugs at both ends of the line. The oversized outer vinyl

jacket was then shrunk in place over the entire length of

line, and over the plugs at the ends, as well, so as to make

them an integral part of the line. A high- temperature heat

gun was used to perform the shrinking.

The completed 30 -foot length of line was tested in a high

level electromagnetic field. The interaction with the field

was found to be approximately two orders of magnitude less

than that found for the copper line used as a reference. A

high noise level was found to exist on the line, due to

changing contact resistance between the carbon particles

(when the line was flexed) . This noise was found to average

approximately 0.2 volt on open circuit, and approximately 0.02

volt when one end of the line was terminated in a resistance

of 10 megohms. However, since the d-c level transmitted over

this line was in the range from 1 to 10 volts, the "flexural"

noise did not present a problem. The Teflon line was found

to be considerably more stable electrically than the other

plastic lines tested, and therefore was adopted for use with

NBS near-zone field-strength meters.

40
7 . 4 Electrical Characteristics of the Conduct ing-Tef Ion Line . Conductor Resis t ance . The nominal volume resistivity of the

conduct ing-Tef Ion plastic material used is 3 ohm-cm. The

measured d-c resistance of this material in . 03-inch-diameter

monofilament form is 20,000 ohms per lineal foot, giving a loop resistance for the twin-conductor balanced line of

40,000 ohms per foot.

Mutual Capacitance . The mutual capacitance between the

parallel conductors of this line, as measured at a low audio

frequency, is approximately 10 picofarads per foot.

Propagation Function . The degree of interaction of the conducting plastic transmission line with the field being measured is directly related to the line loss, or attenuation, of

the resulting wave induced on the line. By making the attenua

tion sufficiently great, this interaction can be reduced to

negligible proportions. The propagation function of this

transmission line can be derived from the classical transmission line equations to be found in any engineering text on

the subject [13]. This theory becomes only approximate in the

case of the lossy line, however, and is offered here only to

illustrate the degree of attenuation to be expected for such

a line. In regard to the electrical parameters of this line,

it can be shown, at least for frequencies below 100 MHz, that

the series inductive reactance per unit length, ojL, is negligible compared to the series resistance per unit length, R,

and that the shunt conductance per unit length, G, is negligible compared to the shunt susceptance per unit length, ooC

.

That is wL << R, and G << ooC . When these conditions are

substituted in the complete equations for the attenuation and

phase functions of the ref lectionless transmission line, the

following expressions are obtained:

41
Attenuation Function a = —*— , nepers per unit length. (19)

~ TwCR 1 h

a = L^J '

Phase Function 3 *[V)\ = —^— , radians per unit length. (20)

It is interesting to note that for this case, both the attenuation and phase functions are given by the same expression. These quantities both increase as the square root of

frequency over the frequency range for which the equations

are valid.

7.5 Transmission Li ne Attenuation . The attenuation per unit

length along the line can be determined approximately from

equation (191 for both the balanced- and the common-modes of

propagation, by substituting in appropriate values of R and C.

The use of the values R =40,000 ohms/foot, and C = 10 pF/foot,

given previously, yields a balanced-mode attenuation of approximately 6.1 nepers (53 dB) per foot at a frequency of 30 MHz

for the conducting-Teflon line. To determine the common -mode

attenuation, the two transmission-line conductors are considered as operating in parallel, making R' = 10,000 ohms/foot.

The value of resistance to use in equation (19) is R = 2R' = 20,000 ohms/foot. If the insulated line is lying flat on the

ground plane, its capacitance, C', to the plane can be considered to be approximately four times that given previously,

or 40 pF/foot (assuming the spacing between conductors to be

double their height above the image plane) . The value of

capacitance to use in equation (19) is then C = C'/2 = 20 pF/

foot. This, then, results in the same r-c product, and therefore the same attenuation for the common-mode case as for the

balanced-mode of propagation.

Thus, it can be seen that the line loss at 30 MHz for

either mode of propagation is extremely high for the conditions assumed. This means that the r-f current flowing at

42
any point along the portion of transmission line, lying on the

ground plane, will be attenuated by more than 40 dB (100X) in

traveling a distance of only one foot. The common-mode attenua

tion decreases somewhat for the short portion of line near theprobe, since its capacitance to the plane is less. However,

this has presented no observable difficulty, as long as this

section of line is maintained normal to the probe axis to preserve the electrical symmetry. From actual tests made on this

line in an electromagnetic field, the attenuation achieved in

this line appears to be sufficient to reduce the interaction

with the field to negligible proportions.

The attenuation was measured in the laboratory from d-c

to 100 MHz using a conducting-Teflon monofilament as the

center conductor of a 5-foot length of rigid coaxial line

having a copper outer conductor 1.25 inches inside diameter.

The measured attenuation agreed with the theoretical, based

on classical transmission-line theory, to within a proportion

of 1 dB in 10 over the above frequency range. The results

are shown plotted in figure 19.

As can be seen from equations (19) and (20), the phase

shift, in radians, along the line is equal to the attenuation

in nepers for the same distance. It is a little surprising

to find that the 30-foot line used in our tests is one-half

wavelength long at approximately 10 kHz, due to the extremely

rapid phase shift along the line. Conversely, the velocity

of propagation, v, along the line is extremely low, as can be

determined from equation (21), which is based on the relation,

v = o)/3.

v ~s]\ I rw I » unit distance per second. (21)

43
The velocity of propagation for the above line is approximately

0.094 v , at a frequency of 30 MHz, where v = free-space

velocity. The velocity is proportional to (w) 2 over the frequency range for which equation (21) is valid.

44
lOOd 83d SNVIOVd *# 'ld!HS 3SVHd

00 10 *

u

•HM

oo

o3 rHPh

bfl PS •H

+-> u

3

T3

PSO

O

0) ^+->

M-lO

N

+-> z m

2 •H

„ XS

>- CO o

z <D Z) 10 o a3 tu ,£

u. PX

13

PS

03

PSO

•H

4-> o33 • PS <D <D pj

4-> -H

•P rH <

PiO

• •H

a. in i—

i

CO •H

CD e

r-l 10 3 PS

bfl nj •H Jh fc +->

CO <0 4- 100J d3d Sd3d3N '» 'NOIlVflNSllV

45
8. CONCLUSIONS

The near-field probes described in this publication will

enable researchers to accurately measure hazard-level, nonionizing, electric and magnetic fields in conjunction with

their various radiation exposure programs being carried out

in the frequency range from 10 to 1000 MHz. These probes in

conjunction with an electromagnetic near-field synthesizer,

recently developed at NBS [14] , will also enable researchers

to evaluate any near-field effects encountered in connection

with their studies of r-f biological hazards.

The recent discovery at NBS and elsewhere of the leading

importance of the role played by the magnetic component of the

field in the area of r-f biological hazards to humans in this

frequency range, points up the need for accurate magneticfield probes for further evaluating this effect.

The two sizes of dipole- and loop-antenna probes constructed

were originally intended for use at frequencies from 10 to 30

MHz only. If the frequency corrections, resulting from the

performance analysis discussed, are applied, the estimated

measurement uncertainty will be less than 0.5 dB (6 percent)

at those frequencies for which the overall length of the dipole

does not exceed A/8, or the loop diameter, A/100. This will

permit electric-field measurements at frequencies up to 750 MHz,

and magnetic-field measurements at frequencies up to 75 MHz.

The addition of the smaller, third dipole and loop probes,

treated theoretically, will extend these frequencies to 1500

MHz and 250 MHz, respectively.

Thus, it is possible to reduce the measurement uncertainty,

or to extend the usable frequency range, by following the procedures outlined. The probes have performed reliably and provide an electromagnetic hazard-measurement capability not

previously available in this frequency range.

46
9. REFERENCES

[I] Jordan, E.C., Electromagnetic Waves and Radiating Systems,

p. 334 CPrentice-Hall, Inc., Englewood Cliffs, N.J., 1950).

[2] IEEE Standard Methods for Measuring Electromagnetic Field

Strength, IEEE No. 302, eq. (10). [The Institute of

Electrical and Electronics Engineers, Inc., 345 East 47th

Street, New York, N.Y. 10017, August 1969).

[3] Jordan, E.C., op. cit., pp. 333-338.

[4] Schelkunoff, S.A., and Friis, H.T., Antennas, Theory and

Practice, p. 415, eq. 43 (John Wiley $ Sons, Inc.,

New York, N.Y., 1952)

.

[5] Greene, F.M., NBS Field-Strength Standards and Measurements, Proc. IEEE, Vol. _55, No. 6, pp. 970-981 (June 1967).

[6] Ramo, S., Whinnery, J.R., and Van Duzer, T. , Fields and

Waves in Communication Electronics, pp. 228-242 (John

Wiley § Sons, Inc., New York, N.Y., 1965).

[7] Kraus, J.D., Antennas, pp. 155-172 (McGraw-Hill Book Co.,

Inc., New York, N.Y., 1950).

[8] Whiteside, H., and King, R.W.P., The Loop Antenna as a Probe, IEEE Trans, on Antennas and Propagation, Vol. AP - 1

2

, No. 3, pp. 291-297, May 1964.

[9] Whiteside, H. , and King, R.W.P., loc. cit.

[10] Whiteside, H. , and King, R.W.P., loc. cit.

[II] Terman, F.E., Electronic and Radio Engineering, pp. 44-57

(McGraw-Hill Book Co., Inc., New York, N.Y., 1955).

[12] Greene, F.M., A New Near-Zone Electric-Field-Strength

Meter, NBS J. of Res., Vol. 71-C (Eng . and Instr.), No. 1,

pp. 51-57 (Jan. -Mar. 1967).

[13] Guillemin, E.A., Communication Networks, Vol. II, p. 69

(John Wiley $ Sons, Inc., New York, N.Y., 1935).

[14] Greene, F.M., Development and Construction of an Electromagnetic Near-Field Synthesizer, NBS Technical Note 652

(May 1974). For sale by the Superintendent of Documents,

U.S. Government Printing Office, Washington, D.C., 20402

(Order by SD Catalog No. C13. 46:652) $0.50.

47

U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET

1. PUBLICATION OR REPORT NO.

NBS TN-658

2. Gov't Accession

No.

4. TITLE AND SUBTITLE

Development of Electric and Magnetic Near-Field Probes

3. Recipient's Accession No.

5. Publication Date January 1975

6. Performing Organization Code

7. AUTHOR(S)

Frank M. Greene

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

10. Project/Task/Work Unit No. 2767433

11. Contract/Grant No.USAF

Memorandum of Agreement dated 7/1/73

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

USAF School of Aerospace Medicine,

Brooks Air Force Base, Texas 78235

13. Type of Report & Period

Covered Finai Report

7/1/73 to 6/30/74

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here.)

This publication describes the development and design of small electric and magnetic near-field probes for measuring hazard-level fields up to 20,000 V/m and

100 A/m, respectively. They were originally designed to be used over the frequency

range from 10 to 30 MHz, and consist of short dipole antennas and small, single- turn

balanced, loop antennas to measure the electric-and magnetic-field components, respectively. The probes are intended for use by various researchers in their electromagnetic, radiation-exposure programs for determining the effects of hazard-level,

non-ionizing, EM fields on living tissue, electro-explosive devices, and volatile fuels.

In order to later extend the use of the probes to frequencies above 30 MHz, a detailed analysis was made of several types of measurement errors likely to be encountered. The principal errors result from a variation with frequency in: the effective

length and impedance of the dipoles; and the electric-dipole response and partial resonance of the loops. Corresponding corrections are given for each type of error as a function of the operating frequency from 10 to 1000 MHz, and as a function of the

physical and electrical sizes of the probes.

As a result of the analysis, the dipoles can now be used for measurements at frequencies up to 750 MHz and the loops to 75 MHz with an estimated uncertainty of 0.5 dB.

Applying recommended corrections will provide a substantial further increase in the

usable frequency range.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons)

Electric near-field probe; electromagnetic-field hazard; field-strength measurements;

magnetic near-field probe; near-field measurements; r-f hazard measurements;

semiconducting transmission line.

18. AVAILABILITY |x Unlimited

| For Official Distribution. Do Not Release to NTIS

I

' Order From Sup. of Doc, U.S. Government Printing Office Washington, D.C. 20402, SD Cat. No. C13.46: 658

| ! Order From National Technical Information Service (NTIS)

Springfield, Virginia 22151

19. SECURITY CLASS

(THIS REPORT)

UNCLASSIFIED

20. SECURITY CLASS

(THIS PAGE)

UNCLASSIFIED

21. NO. OF PAGES

53

22. Price

$1.10

USCOMM-DC 29042-P74

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