*• W ^ -fc NBS TECHNICAL NOTE 667

U.S. DEPARTMENT OF COMMERCE / National Bureau of Standards

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Upper-Bound Errors in

Far-Field Antenna Parameters

Determined From

Planar Near-Field Measurements

PART 1: Analysis
NATIONAL BUREAU OF STANDARDS

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Upper-Bound Errors in

Far-Field Antenna Parameters

Determined From

Planar Near-Field Measurements

PART 1: Analysis

A.D. Yaghjian

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, CO 80302

Sponsored by

Air Force Avionics Laboratory

Air Force Wright Aeronautical Laboratories

Air Force Systems Command

Wright-Patterson Air Force Base, Ohio 45433

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DEC 3,

1 1975

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U.S. DEPARTMENT OF COMMERCE, Rogers C. B. Morton, Secretary

James A. Baker, III, Under Secretary

Dr. Betsy Ancker-Johnson, Assistant Secretary for Science and Technology

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Acting Director

Issued October 1975
NATIONAL BUREAU OF STANDARDS TECHNICAL NOTE 667

Nat. Bur. Stand. (U.S.), Tech Note 667, 120 pages (Oct. 1975)

CODEN: NBTNAE

For sale by the Superintendent of Documents, U.S. Government Printing Office , Washington, DC. 20402

(Order by SD Catalog No. C13. 46:667) $1.85
CONTENTS

Page

List of Figures and Tables iv

I. Introduction 1

CI. Relationship of Errors in Gain Function, Sidelobe Level,

Polarization Ratio, and Beamwidth to the Far-Field Error

in Electric Field 4

II. Error Analysis 8

A. Finite Scan Errors 8

1. Mathematical Formulation of the Problem 9

2. The Fields of Electrically Large Aperture

Antennas 13

3. Evaluation of n(r) 18

B. Position and Instrumentation Errors 30

1. Position Errors 33

2. Instrumentation Errors 48

C. Multiple Reflections 58

IV. Summary 69

Appendix A -- Near-Fields of a Circular Antenna with

Uniform Aperture Distribution 96

Appendix B -- Position and Instrumentation Errors for the

Null Depth of Difference Patterns 105

References 111

Acknowledgment 113

in
LIST OF FIGURES AND TABLES

Page

Figure 1. Main beam of a hypothetical test antenna 82

Figure 2. Schematic of scanning geometry 82

Figure 3. Schematic of aperture antenna 83

Figure to 4. Definition of a , 9 , D 83

m ' m ' m

Figure 5. Schematic of aperture and scan areas 84

Figure 6. Circular reflector antenna 84

Figure 7a. Near-field centerline data (constrained lens)

(z = 25 cm) 85

Figure 7b. Near-field centerline data (reflector antenna)

(z = 43.18 cm) 85

Figure 8a. Change in gain vs. decreasing scan length

(constrained lens) 86

Figure 8b. Change in gain vs. decreasing scan length

(reflector antenna) 86

Figure 9a. Constrained lens sum port near-field log amplitude, f = 9.2 GHz, z = 25.0 cm, no radome 87

Figure 9b. Near-field phase, constrained lens sum port,

f = 9.2 GHz, z = 25 cm 87

Figure 9c. Constrained lens antenna and probe 87

Figure 9d. Near-Field Centerline Data, z = 25 cm 88

Figure 10. Position errors in on-axis gain 89

Figure 11. Comparison with Rodrigue et al . for random phase

errors 89

Figure 12. Comparison with Newell for quadratic phase errors 90

Figure 13. Amplitude errors in on-axis gain 90

Figure 14. On-axis error in far-field from multiple

reflections 91

Figure 15. Far-field errors from multiple reflections

(a/A = 12) 91

IV
LIST OF FIGURES AND TABLES (continued)

Page

Figure Al . Definition of t and a 102

Figure A2. Dotted line shows region in which eq . (A12a)

holds 102

Figure A3. Near-field amplitude of circular antenna 103

Figure A4 . Near-field phase of circular antenna (a/A = 12)- 104

Table 1. Finite Scan Errors from Centerline Data of a Typical X-Band Antenna 92

Table 2. XY- and Z-Position Errors for a Typical X- and

K-Band Antenna 9 3

Table 3. Instrumentation Errors in Measuring Amplitude 94

Table 4. Total Far-Field Errors for a Typical X- and

K-Band Antenna 9 5

v
UPPER-BOUND ERRORS IN FAR- FIELD ANTENNA PARAMETERS

DETERMINED FROM PLANAR NEAR- FIELD MEASUREMENTS

Part I : Analysis

A.D. Yaghj ian

ABSTRACT

General expressions are derived for estimating the errors in the

sum or difference far-field pattern of electrically large aperture

antennas which are measured by the planar near-field scanning technique. Upper bounds are determined for the far-field errors produced by 1) the nonzero fields outside the finite scan area, 2) the

inaccuracies in the positioning of the probe, 3) the distortion and

nonlinearit ies of the instrumentation which measures the amplitude

and phase of the probe output, and 4) the multiple reflections.

Computational errors, uncertainties in the receiving characteristics

of the probe, and errors involved with measuring the input power to

the test antenna are briefly discussed.

Key words: Antennas; error analysis; far-field pattern; near-field

measurements; planar scanning; planewave spectrum.

vr
I . Introduction

The planar near-field scanning method has been used to measure

with high accuracy the electromagnetic fields of microwave antennas.

High accuracy is possible primarily because so few restrictive

approximations are involved in the formulation and application of

the near-field techniques. Moreover, the far-field errors associated

with each approximation can be estimated because the approximations

themselves can be expressed in convenient mathematical form. This

report essentially derives and evaluates some of these mathematical

expressions under the given laboratory conditions to determine quantitatively the accuracy with which the far-field of antennas can be

measured by the planar near-field scanning method.

The error analysis was undertaken for two main reasons. One,

it was desired to find general upper-bound expressions for the

limits of accuracy in computing far-fields from planar near-field

measurements without resorting to direct far-field comparisons. It

has long been the feeling of those involved with the near-field

measurement techniques at the NBS that these techniques were often

more accurate than measurements taken on conventional "far- field"

ranges with a standard antenna. Thus comparisons with patterns

measured on conventional far-field ranges would not give a reliable evaluation of the near-field techniques which do not have to

cope with proximity corrections, ground reflections, or the calibration of standard far-field antennas. Two, the upper-bound expressions could be used to stipulate design criteria for the construction of new near-field scanning facilities. This meant that

the upper-bounds for the accuracy in a given far-field parameter

should be expressed in terms of measured near-field data and/or the

computed far-field, the frequency and dimensions of the antenna-probe

system, the variation in the positioning of the scanner, and the

precision of the instrumentation which measures the probe output.

The design engineer could then compute from the upper-bound expressions the near-field tolerances required to insure a given farfield accuracy for the range in size and frequency of the antennas

he was considering.

There have been two major computer studies performed in the

past to estimate the errors involved with planar near-field scanning
measurements [11,16]. Rodrigue, Joy and Burns [16] have introduced

position and instrumentation errors into a hypothetical near-field

distribution in order to compute their effects on the far-field.

Newell and Crawford [11,17] performed a similar analysis with measured near- field data which included an estimate of errors involved

with truncating the scan plane. (Jensen [25] has also made a numerical

investigation into the accuracy of near-field measurement techniques

but for nonplanar scanning only.) A major drawback of the computer

studies, as well as direct far-field comparisons, is that they

apply to particular antennas and their results do not necessarily

represent upper-bounds to errors which will hold for large general

classes of antennas. It should be emphasized, however, that the

computer studies are an extremely useful aid in giving direction to

the general analysis and checking its results.

This report does not attempt to estimate the accuracy of the

extrapolation technique for measuring the gain of antennas [22]

.

An error analysis involving the extrapolation technique has been

performed recently by Kanda [23].

If we assume that the antennas under consideration are linear,

of finite extent, and operating in a single mode at fixed frequency

and amplitude, and that Maxwell's equations for free space describe

the region in which the antenna is situated, the only approximations involved in formulating and applying the planar scanning

method are

:

1) The fields outside the finite scan area are zero.

2) The scanner is aligned and positioned with infinite

precision

.

3) The instrumentation introduces no distortion and measures

the amplitude and phase of the probe output with perfect

accuracy

.

4) Multiple reflections between the test antenna and "probe"

antenna are zero

.

5) Computation errors in "deconvolut ing" the measured nearfield data to get the far-field are nil.

Errors caused by uncertainties in the receiving characteristics of

the probe, and errors involved with measuring the input power to the

test antenna are discussed in Section IV, which, incidently, contains

a summary of the major results and conclusions of this report.
The far- field errors introduced by the computations (approximation 5) are so much smaller than the combined errors in the farfield caused by a finite scan area, positioning, instrumentation,

and multiple reflections (approximations 1-4) that they are of

little consequence. With the help of the sampling theorem, Fast

Fourier Transform, and computers accurate to many places, the

necessary deconvolution of the measured data can be performed with

insignificant error.

The amplitude of the near-field multiple reflections (approximation 4) can be estimated in practice by changing the distance between the probe and test antennas. Any periodic variations in the

amplitude of the received signal repeating about every A/2 (A = wavelength) would be caused primarily by the multiple reflections.

In Section III.C the effects on the far-field of multiple reflections

are discussed and estimated analytically. Although they cannot be

eliminated completely, multiple reflections can be reduced by using

efficient absorber material, by decreasing the size of the probe, or

by increasing the distance between the probe and test antenna.

Also, it is likely that the effect of multiple reflections on the

far-field can be reduced by averaging the far-field patterns obtained by scanning on a number of near- field planes which are

separated by a small fraction of a wavelength.

Far-field errors caused by the errors in the scanner positioning and instrumentation (approximations 2 and 3) are determined from

a common set of equations. These equations are derived and evaluated in Section III.B for both systematic and random near-field

Without the advantage of the FFT, typically the computer would add

20,000 terms to calculate the variable which yields the far-field.

Even if we make the absurd hypothesis that every round-off error

adds in the same direction, a computer of 10 -place accuracy adding

20,000 numbers would retain more than 5-place accuracy (.001%) for

the sum. Of course, in practice the FFT algorithm vastly reduces

the number of necessary computations.

There is also the approximation associated with applying the sampling theorem, which assumes that the output of the probe is the

Fourier transform of a band-limited function. For a separation distance between test antenna and probe of more than a few wavelengths,

this can be shown to be an extremely good approximation which introduces negligible errors into the far-field.
errors. "Position errors" may be reduced by scanning along both

vertical and horizontal lines and appropriately averaging the two

sets of measurements. "Instrumentation errors" can be reduced if

the distortion and nonlinearit ies of the receivers can be determined

and included as part of the computer program that deconvolutes the

near-field data.

The first portion of the error analysis (Section III. A) is

devoted to determining the maximum far- field errors introduced by

neglecting the fields outside the scan area. As part of the analysis, asymptotic expressions are derived for the near-field in front

of large aperture radiators . All errors are assumed small enough so that the individual contribution to the far-field error from each source of near-field

error can be estimated independently and then combined to give the

total far-field error. We shall find that all the individual errors

in the far- field combine linearly except for the on-axis sum pattern

position and instrumentation phase errors which combine quadratically (see footnote 12).

1 1 . Relationship of Errors in Gain Function, Sidelobe Level,

Polarization Ratio, and Beamwidth to the Far-Field Error in

Electric Field

Let E (r ) denote the electric field (to within the limits of

error) of an antenna radiating into free-space and ± AE(r) the

limits of error involved with the measurement of E(r) (e time

dependence has been suppressed). The magnitude of the fractional

error n(r) in the electric field amplitude can be defined for small

errors as

,

n(r) = E±AE AE(r)

|E(F)

(1)

(Henceforth, the < sign will be omitted when eq . (1) or similar expressions are used.) The Hermitian amplitude |E| = /E # E* (the

asterisk * denotes the complex conjugate) is related to the power S

radiated per unit area in the far-field or any other locally
plane-wave field in free-space by S = /e /y | E

|

2 From the values

of n(r) as r approaches infinity and the approximate amplitude of

the far-field |E(r)| , all other errors in far-field quantities

can be found. Four such quantities of interest are the gain function, the sidelobe level, the polarization ratio, and the half-power

beamwidth. Of course, these four are not always the only far-field

quantities of interest, but they will be used as typical examples

to demonstrate how the error in any far- field quantity can be related easily to n(r) and |E(r)| . (In eqs . (2)-(5) below, where

the error in these four quantities is derived, the subscript M r+°°"

is understood, but not shown explicitly.)

The error n^ in the gain function G(r), which is proportional

to the square of the far-field amplitude, may be written immediately

as (for small |AE~|/|e|, so that |AE| 2 terms can be neglected)

E ± AE El 2

1 I AE

,

G = ± 2 U±Ba. g (2a)

or in decibels

= ±2n(r)G(r),

dB _ -, n -. n G

= 10 log E ± AE 2 ^ = 20 log(l±n(r)) (2b)

For small errors, n << 1 , and

(2c)

The sidelobe level, when meaningful, is usually stated in deci

bels and is defined as the ratio of the maximum far-field intensity

of the largest sidelobe to the maximum far-field intensity of the

main beam [1]

For SL = 'side

The error r\ in sidelobe level SL is

s

'side ± AE side side

E ± AE

max max

E | 2 max '

VIEmax

and small r\ (r) ,

r, s = ±2[n(r

s

) + n(r

Q )]SL, (3a)
with n(r ) and n(r ) denoting the values of n at the sidelobe maxi

mum and main beam maximum respectively. In most cases n(r ) will

be much larger than n (r ) so that r\ = ±2ri(r )SL. (3b)

In decibels

dB

n = 20 log side AE

s ide

E ± AE

max max

log 'side

max

or for small n , and n (r ) >> n (r ),

dB

7 n(r

s

) (3c)

i.e., as we might expect, simply the error in the gain of the sidelobe itself.

The polarization ratio (Pol) of the electric field is defined as the ratio of the minor to major axis of its polarization ellipse [1]

.

For a given amplitude of error field | AE | the maximum error in the

polarization ratio at a point in the far-field occurs when the field

AE has the same polarization ratio as E but with major and minor axes

interchanged and the direction of rotation of the electric-field

vector reversed.* (Also the electric field vectors of AE and E must

be colinear along the major-minor axes.)

Under these conditions, the maximum error (APol) in polarization

ratio is realized and can be expressed for small errors in the form

APol = ±[1 + (Pol) (4a) l

] n(r)

.

The upper-bound error in polarization is never greater than 2n (circularly polarized) and for antennas with small cross polarization

the upper-bound error is approximately n

.

It is an interesting result that if the maximum error in polarization ratio were realized, the rotational shift in the polarization

ellipse caused by the error field would be a minimum, i.e., zero.

Fortunately, as part of the derivation of eq. (4a) the maximum

possible r rotational shift \br in the polarization ellipse for a max r r given | A E | can also be determined. Specifically,

4> max

1 + (Pol) 2

'

(PoI) 2

J radians (4b)

For a far-field of approximately circular polarization (Pol - 1) the

shift in polarization ellipse introduced by the error field may

*This result, the proof of which is straightforward but rather lengthy,

was obtained by Flemming Jensen and brought to my attention by Flemming

Larsen both of the Technical University of Denmark, Lyngby, Denmark.
become large --as one might expect. Finally, it is mentioned that

footnote 11 should be consulted when applying eqs . (4) to instrumentation amplitude errors.

The half-power beamwidth is defined for a plane containing the

line along the maximum intensity of the beam. It is simply "the

angle between the two directions in which the radiation intensity is

one half the maximum value of the beam" [1] . If we know E(r) and

AE(r) in the far- field we can determine the maximum errors in measuring the beamwidth. In figure 1 and |E ± AE 2 E AE

(neglecting | AE

|

2 terms, as usual) are plotted for the main beam of

an arbitrary test antenna. The beamwidth is 0,+6

2

. From figure 1,

we see that the error in 9-. is

A0

1 = ± 2|AE(9

1

)| |E(6

1

) AE(6

1 )| |E(8

1

) tan a E(9

1

) d|E(9

1

)

±n(9 1 )|E(9

1

) dlECe^l

radians . **

~dQ d9

A corresponding expression holds for the error in A9~. The total

fractional error n R in beamwidth may be written

(5a)

(5b)

or, for symmetrical beams (9=9,= 9~), simply

n B

= ±A9

9

. ±ri(9) E(9)

fi d|E(9)|

d9

For large circular or square apertures of uniform amplitude and

phase distribution, |F(0)|/ e^-L

equal 1, and nTM becomes simply,

phase distribution, |E(0)|/ 9

L-t4 j [ can be shown to approximately

! B :n(8) (5c)

As expected, eqs. (2)- (5) verify that a knowledge of |E(r)|

and n(r) as r -» °° (as mentioned above, the "r > °°" is suppressed in

eqs. (2) -(5)) is all that is required to determine the errors in the

gain function, sidelobe level, polarization ratio, and beamwidth.

If desired, errors in other far-field parameters can equally well

be expressed in terms of |E(r)| and n (r) . The far-field |F| may

be computed from the measured near-field data or estimated analytically. Thus, the problem of finding the far-field errors reduces

to the problem of evaluating n (r ) of eq. (1) in the far-field.

**Here we are neglecting the error caused by the change in the_maximum

value of the beam. If this error is significant compared to AE(9,) it

should be added to it .
III. Error Analysis

A. Finite Scan Errors

The purpose of the present section is to estimate the maximum

errors in the far-field introduced by scanning in the near-field

over a plane of finite area. "Finite" is the key word. In principle, the planar scan method requires that the output of the probe

be recorded over an infinite plane in front of the test antenna.

In practice, of course, only a finite area of the plane is scanned

and the fields outside that finite area are set equal to zero. The

scan area is usually, but not necessarily, rectangular with the

boundaries commonly chosen where the output of the probe antenna is

down 30 or 40 dB or more from its maximum. For many microwave antennas such a scan area turns out to be about two or three times

the aperture area of the antenna. The present analysis will be restricted to electrically large (average width/wavelength > 10 will

do) aperture-type antennas, usually but not necessarily operating

at microwave frequencies. The phase will be assumed fairly uniform

across the aperture with the amplitude of the field either uniform

or reaching a maximum near the center of the aperture and convexly

2 tapered toward the edge to reduce the sidelobe radiation [2].

Near-field, centerline, amplitude and phase data for the sum pattern

of a typical antenna measured at NBS can be seen in figure 9d . Reflectors, large horns, and broadside arrays are probably the most

common examples of the test antennas under consideration. The results of this finite scan part of the error analysis apply to

antennas with their boresight direction steered at an arbitrary

angle with respect to the scan plane.

Initially we are assuming the antenna is operating in a sum mode or

pattern. For finite scan errors it will be shown that difference

patterns need not be considered separately since they are formed by

the superposition of two sum patterns with wavefronts slightly

skewed to create a "null" in the boresight direction. For

position and instrumentation errors, however, Section III.B along

with Appendix B shows that sum and difference patterns must be considered separately, at least when determining far-field errors near

the boresight direction.
The approach which is used to estimate the finite scan errors

involves finding an upper-bound to the appropriate integral of the

fields outside the finite scan area. Physical optics and the geometrical theory of diffraction are used to show that the fields outside the scan area are determined chiefly by edge diffracted fields

of the antenna. For each antenna these edge diffracted fields are

different. However, they all can be expressed in a general form

(eq. (19)) which allows upper-bound expressions (eqs. (32) and (36))

to be found by evaluating the integral outside the scan area in terms

of the probe output on the edge of the scan area. Outside the

"solid angle" formed by the edge of the aperture and edge of the

scan area, the evaluation of the integral can be done by the method

as stationary phase to show that in this region the far-fields computed from the near-field data cannot be relied upon with any confidence. Well within the solid angle the integral evaluation can

be performed through integration by parts to yield an upper-bound

expression for finite scan errors from both centerline data scans

(eq. (32)) and full scans (eq. (36) or (32)).

1 . Mathematical Formulation of the Problem

Suppose we want to determine the radiation pattern of a given

antenna of aperture area A bounded by the curve C, as in figure 2.

The task is accomplished experimentally by recording the output of

an arbitrary but known probe antenna (for two orientations, in

general) as the probe scans in front of the radiating test antenna

on a plane of area A' bounded by C

1 . The z-axis will always be

chosen perpendicular to the scan plane with origin in the antenna

aperture. The scan plane, however, may not always be chosen parallel

to the aperture plane. For beams steered off -axis, larger scan areas

may be required if the scan plane does not lie perpendicular (approximately) to the boresight direction.

After taking a double Fourier transform of the probe output in

each orientation, the radiating characteristics (S, (K) ) of the test

antenna are found simply by solving simultaneously the two resulting

linear equations. In particular, if the probe were a perfect dipole,
the output of the probe would be proportional to the electric field

at the dipole in the direction of the dipole moment. The expression

for S, (K) (defined with respect to the reference plane z = 0, and

the transverse part of the propagation vector denoted by K) then

becomes [3]

S l0 (K) = —7— e iYd

/ E (P,d)e" iK ' P dP, (6)

o

f - ioot j , . (e time dependence)

where E (P,d) is the electric field, i.e., the output of the dipole

probe in two mutually perpendicular orientations, transverse to the

z-axis at the point (P,d) in the scan plane A'. (The equation of

the A' plane is z = d.) The amplitude of the input mode to the test

antenna is designated by a , and the variable y is defined by

1, 7 TT Y

= (k 2 -K 2 )'2 (k = 00/c = —r) , where the radical is chosen to keep y

positive real or imaginary. ' For the sake of simplifying the theory we will assume a perfect

electric dipole as the probe. The dipole gives information about

the electric field only at a point. All physically realizable probes

respond to a weighted average of the field near the probe. Thus it

is expected that the errors in the computed far-field introduced by

omitting a part of the infinite scan plane are as great or greater

for a perfect dipole than for any other probe antenna, and that the

following conclusions and resulting upper bound expressions (32) and

(36) hold for arbitrary probes. Also, at this point in the error

3a The sampling theorem shows that to obtain the far-field pattern

the double integral in eq . (6) can essentially be replaced by a double summation over points in A' separated by about A/2 or less.

Thus, in practice, data need be taken only at a finite number of

discrete points for eq . (6) to be evaluated. However, for most of

the error analysis_we prefer (somewhat arbitrarily) the integral

representation r of S. to the summation. lo

Strictly speaking, the Fourier transform in eq. (6) and eqs . (7),

(10), and (11) below may not converge^ to a unique value as the scan

area approaches infinity because E+-(P,d) has a 1/P dependence in a lossless medium as P->°°, and thus gives rise to a rapidly oscillating

part in the transform as P-*». Usually this oscillatory term can be

ignored with impunity because it vanishes upon integration when

taking the inverse transform. However, it does determine the limiting value of the finite scan errors (see footnote 9).

10
analysis any uncertainty in the receiving characteristics (S') of

the measuring probe are ignored.

For eq. (6) to represent S, (K) exactly, the probe scan and

subsequent integration would have to be performed over the infinite

plane. r Thus the error (AS, * ) produced in S, by a scan of finite lo^ r lo J area may be written formally as

AS- = —^—

e"

iYd

/ E (P,d)e~ iK ' P dP, (7) 10 4^ 2 a A'

t o c where A

1 is the infinite area outside or complementary to A' . The

far-fields can be found from S, (K) by taking the inverse double

Fourier transform of eq . (6) and evaluating the resulting expression

by the method of stationary phase for double integrals. So doing

yields [3]

ikr

E.(r) = -27ia ik cosB S, (kR/r) (8) ^

v j lo v J r

X" -> co

and from eq. (8)

ikr

AE^(r) = -2iTa ik cosGAS, (kR/r) . (9) t

K J o lo v r

r -> oo (cos9 = z/r, z>0)

Substitution of S, and AS, from eqs. (61 and (7) into eqs. (8) and

lO lO ivy v. j -i v j (9), respectively, produces expressions for the far -electric-field

and its error in terms of the near-field data:

E (7) = - ik C056e ik(r - d C0S ^/ E

t (P,d)e"

lFR ' P

dP (10)

Z

27rr A'

r

X" -*- °o

AEJ7) = -i\^- e ik(r " d COs9)

/ E

t

(P,d)e"

lFR ' P

dP. (11)

c

X* -> oo

(For a dipole probe, E (P,d) and the output of the probe are identical.) Equations (10) and (11) express mathematically the well-known

11
"Fourier optics" result that the far-field amplitude is proportional

to the spatial Fourier transform of the near-field times cos0.

Division of eq . (10) by eq. (11) results in the fractional error r\ in the transverse part of the far-electric -field

n

t

(r)

AE

t

(r)

E

t (r)|

-i-R-P

/ E.(P,d)e r dP

A'

z

-i-R-P

| E (P,d)e r dP

A'

Z

(12a)

or simply

cos6

n t

(r) =

-i- R-P

{ E. (P,d)e r dP

A'

z

__cAr |1\ (r) | (12b)

Of course, the fractional error r\ in the total far-electric

field is not necessarily n f except on the z-axis. However we can

find n in terms of n

f by a simple argument. In the far-field |E (r)

differs from |E(r)| at most by a factor cosG, i.e.

|E(r) |cos6 _< |E (r) | < |E(r)

Similarly

AElcosG < AE

t

| < AE

(12c)

(12d)

Consequently ri is greater than n t by at most a factor

Q

, and from

eqs . (12) we can express the maximum possible n in the far-field as

n(r) = AE

/ E.(P,d)

A'

Z

-i- R-P

e

r dP

Xr|E(r)| (13)

•]"->• 00

For the denominator of eq . (13) we can use the far-field estimated

analytically or computed from the measured near-field data. Thus,

the problem of finding n reduces to that of estimating E (P,d) on A^,

i.e., on the area outside the finite scan plane.

12
2 . The Fields of Electrically Large Aperture Antennas

If the behavior of the electric field E (P,d) were known in the

area A*, the integral of eq. (13),

r

I = / E.(P,d")e dP, (14)

A t L A

c could be evaluated and the far-field errors could be found immediately

from eqs . (l)-(5). Even asymptotic methods cannot be applied to eq

.

(14) until the behavior of E (P,d) is determined. Fortunately, the

electric field outside the aperture of electrically large aperture

antennas can be determined analytically from the electric field distribution across the aperture. In fact, it will be shown shortly

that just the electric field at the edge of the aperture distribution

is required.

As a first step in finding the electric field E (P,d) to use in

eq. (14), consider the aperture antenna drawn schematically in

figure 3. (The boundary or rim of the antenna is assumed to lie

in a plane with e here chosen perpendicular to the plane. These

restrictions are relaxed later.) It is well-known [4] that the

electric field everywhere to the right of the infinite plane A^ can

be expressed in terms of an integral of the electric field over A^,

EOO = - ^

/ [e

z xEt

(R')] x VG(F,R») dR' , (15)

00 where

lklr-R' I G = SL-! ' |r-R»|

and E. (R' ) is the transverse electric field on the plane A^ emanating from sources to the left of A^ (e.g., electric fields from a feed located to the right of A^ would not be included in E.(R'),

except, of course, indirectly as reflected fields from the antenna).

For an aperture antenna, whether it be a reflector, large horn,

or broadside array, the components of the electric field E. (R T ) are

13
4 slowly varying in phase and amplitude across the aperture, except

possibly right near the edge. The amplitude often tapers toward the

edge of the aperture and drops abruptly (in a distance less than a wavelength) to zero or near zero beyond the edge. Thus the limits

of integration in eq. (15) reduces to the aperture area A or at most

a couple of wavelengths beyond A. Also, since eq. (14) requires

only the transverse component of E (r) , we can ignore the z- components of eq. (15) and write

E

t

(r) = ' 17 |y / E

t

(R')G(F,R') dR'

.

(16)

For electrically large apertures, eq . (16) can be evaluated asymptotically. Before doing this a couple of remarks are in order.

The first has to do with neglecting the fields outside the

aperture area A when in reality there may be scattering from the

edge of the antenna aperture. In those cases it may seem unreasonable

to neglect, initially, the fields beyond the aperture area A as part

of the procedure to calculate the fields beyond the scan area A'

.

The "canonical" problems that can be solved exactly, such as scattering by an infinite wedge or elliptical disk, indicate, however, that

at short wavelengths the scattered fields are caused predominantly

by high intensity fields within a wavelength or so of the edge, and

indeed the fields more than this distance beyond the edge may be

neglected [5]. Of course, this assumption has been confirmed by

experiment and constitutes the basic postulate of the geometrical

theory of diffraction [6]. In terms of the radiation from large

aperture antennas, it means simply that the fields everywhere to

the right of the aperture plane depend solely upon the fields within

and near the edge of the aperture, even when appreciable scattering

from the edges is present.

The second remark concerns antennas that have part of their

feeding mechanism mounted to the right of the aperture plane--as

in the case of reflector antennas. As mentioned above, the direct

radiation from the feed must not be included as part of E.(R') for

eq . (16) to be correct. However, scattering from feed mounts

4For electrically steered arrays the planes of "uniform" phase may

be skewed with respect to the aperture plane. This situation is

considered at the end of the section.

14
(including the feed itself) of the fields reflected by the antenna

are not taken into account by eq . (16). Although the mount scattered

fields can be ignored if they are small compared to the fields

scattered from the aperture edge, there is no reason to believe this

will always be the case even in the region A'. Fortunately, the

results of the geometrical theory of diffraction (GTD) can also be

used to determine the behavior of the fields diffracted from the

mounting in front of the aperture as well as from any sharp edges

at the boundary of the aperture [6]. Moreover, we shall find that

the final expression (eqs. (32) and (36)) for the far-field error

does not require an explicit estimation of either the mount or edge

diffracted fields. Of course, for many antennas such as horns and

arrays there are no obstructions in front of the aperture.

For the moment we shall ignore the problems of the exact nature

of the edge diffraction and diffraction from feed mounts and return

to evaluate eq. (16) for a smoothly tapered amplitude within A and

zero outside A. For apertures which are many wavelengths across,

the double integral (16) can be expanded in an asymptotic series.

The first three terms in the series for integrals like eq. (16) have

been derived by Van Kampen [7]. Keller, Lewis and Seckler [8] apply

Van Kampen' s results to eq. (16) specifically. The final expression

has been confirmed by the present author using an approach different

from Van Kampen' s. In the shadow region, i.e., the entire half space

z > excluding the cylindrical volume formed by the projection of

the aperture area A along the z-axis, the electric field in eq. (16)

is approximated by

/

E\ (F) = I \- t

v J L 2tt m

am

amm+D sin6 W

H cos9m E.

m tm

e sin0

m

m 4

(17)

The variables in eq. (17) are defined with the help of figure 4.

The distance from the point r to the edge of an aperture with

a smooth boundary has at least one relative maximum and one relative

minimum. The subscript m simply refers to the mth relative extremum

point on the edge° of the aperture. r D is the distance from r to the

m

mth relative extremum,7 a is the radius of curvature of the edge (in

m

b K

15
the plane of the aperture) at the mth relative extremum, 9 is the _ m

angle between D and the z-direction, and E is the value of the

transverse electric field at the edge of the aperture excluding the

fields diffracted from the edge. b The radius of curvature a is taken

m

positive if the distance D is a relative minimum and negative if a

am 1 1/2

m relative maximum. The radical

amm+D sin0

m

is taken positive if

real and negative if imaginary. Of course, all subscripted quantities are, in general, functions of position r. Actually, in order for expression (17) to remain valid, must

be greater than a few A/£ (I e /A) , and D no closer than a couple of wavelengths X from the edge. This latter restriction says

simply that eq. (17) does not describe the reactive fields. The

former restriction can be understood physically by dividing the

aperture into Fresnel zones for the direction . For greater

ave

mm6

than a few X/l there are enough Fresnel zones (even for the farfield) to assure that the main contribution to the fields in the

shadow region are from the fields near the edge of the aperture

rather than from the fields well within the aperture.

The expression (17) does not necessarily account correctly for

scattering from edges that may be present on the boundary or rim of

the antenna. The necessary modification of eq . (17), for example,

when scattering from sharp conducting edges occurs can be extracted

from Kellers GTD [6]. Kouyoumj ian [9a] has written Keller's results

in a form similar to eq. (17). The GTD expression differs from eq.

(17) v J onlyJ in that the factor cos0 E. is replaced by m tm r J

l' 1 g (0 ) + e}- g. (0 ), (18) tm 5II v nr tm 6

_L

V nr ' K J

—II —J- where El and E^ are the transverse components of the incident

tm tm r electric field parallel and perpendicular to the edge of the aperture. The factors g and g can be found from either reference [6a]

or [9a] , but it is not necessary to know them explicitly for our

purposes. Moreover, scattering from other than sharp conducting

edges can also be handled by changing appropriately the factors

& and gj_ in eq. (18)

.

5 Note that in a similar manner it can be argued that the far-fields

for less than a couple X/£max

(&

max = maximum width of aperture)

are determined mainly by the near- fields well within the boundary

of the aperture.

16
Thus it is seen from eqs . (17) and (18) that regardless of the

nature of the scattering from the edges, the fields in the shadow

zone appear to emanate from points along the edge of the aperture.

For our purposes it proves convenient to write simply the one equation valid for the scattered field in the shadow zone

ikDm

E

t

(r) =

I F (r)e m

. (19)

m

F is defined by comparing eq. (19) with eqs. (17) and (18). The

essential property of F (r )

, which allows the asymptotic evaluation

ikD —

of eq. (14), is that unlike e m it varies slowly with r. In fact,

for the error analysis it turns out that this is the only property

of F (r) that is required. Never is it necessary to evaluate F. (r)

explicitly, although for reasons of general interest it has been

evaluated in Appendix A for the circular aperture of uniform distribution. Results from Appendix A are also used in Section III.B.l.

Before carrying out the integration of eq . (14) it should be

mentioned that eqs. (17) and (18) are not valid for large D if the

radius of curvature a approaches infinity. For example, the expressions would be modified if the aperture were rectangular. How- _ _ ikDm

ever, the modifications occur in F (r) but not e . Thus, eq.

(19) remains valid for all shaped apertures even when part of the

edge is a straight line or has infinite radius of curvature.

Also if the edge of the aperture has points where the radius

of curvature is much smaller than a wavelength (e.g., the corners

of a rectangular aperture) , these corners and tips contribute to

the field. For large apertures at least it can be shown [5-9] that

their contribution is usually much smaller (higher order in X/ I ) than the edge fields of eq. (19), and thus can usually be neglected

for the purpose of evaluating eq. (14). However, even if they cannot be neglected, the fields from corners and tips can be expressed

in the same form as eq. (19).

Electronically steered aperture antennas (broadside phased

arrays) are also covered by eq. (19) with the appropriate modifications of F (see, for example, reference [8] which deals with an

arbitrary phase of the field across the aperture) . When the axis

of the main beam is steered away from the perpendicular to the aper17
ture, the region of validity of eq. (19) changes to the entire half

space in the direction of the new axis, excluding the projection of

the aperture along that axis, i.e., eq . (19) is still valid in the

shadow zone regardless of what direction the beam is steered.

Eq. (19) also applies to antennas operating in a difference

pattern where the on-axis field drops to a sharp minimum. As in the

case of the sum pattern, the fields in the shadow zone are determined

by whatever value of electric field impinges upon the edge of the

aperture. And again the variation in electric field separates into

ikDm _

a rapidly oscillating part e and a slowly varying part F . Eq . (17) (or (17) modified by (18)) represents the first term

in an asymptotic expansion of the electric field. The higher order

terms are assumed so much smaller than the first that they are neglected. However, if the electric field at the edge of the aperture

becomes too small, the second term in the asymptotic expansion must

be included. Even then, for reasonably smooth aperture distributions, this second term also has the form of eq . (19) but with F

depending on the derivative of the electric field at the edge rather

than the field itself. This "slope diffraction coefficient" has been

derived for the GTD by Hwang and Kouyoumj ian [9b].

In brief, eq . (19) describes the electric field in the shadow

zone of nearly all large aperture antennas including electronically

steered arrays, antennas excited in a difference pattern, and antennas with aperture distributions that taper to zero at the edge.

3 . Evaluation of n(r)

The evaluation of the integral in eq. (14) and subsequently

n(r) in eq. (13) is accomplished by first substituting the electric

field from eq. (19) into eq. (14),

2tt °° ik[D -p sinBcos (4>-<\> )]

; (V d) e m P

tm L ' J r -' 'P

I =

I J "J F t fP,d)e m P

' J p dp deb. (20)

m

The vectors P and r have been written in cylindrical (p,(b ) and

spherical coordinates (r,0,<b), respectively, defined explicitly in

6 The fields at the edge of the aperture would have to be quite small

for the second term to be significant. For example, an expansion of

eq. (16) for a cosine distribution on a circular aperture of radius

"a" shows that the second term would be required only after the edge

taper became less than about 20 log X/4a (e.g., if a/A = 10, 20 log - X/4a - -32 dB)

.
figure 5. (The z-axis is chosen to intersect the scan area perpendicularly at a point somewhat centrally located.) D' denotes the

distance D when P is on the boundary C of the scan area, which is

always assumed to be outside the aperture boundary C.

Eq. (20) is amenable to the method of stationary phase for

double integrals. The critical points of the first kind of

[D - p sine cos(4>-<f) )] occur at

3D ^ = p sine sin (cf) -cf)) (21a)

^p p

3D

3p^ = sine cos(cj)-cj) ) . (21b)

3D

Consider the derivative ^

, which can be interpreted physically by

referring to figure 5. Let the vector P be the perpendicular proJjection of the line D such that

m

Then

p = /p 2 -Ji 2

. *m r m

3Dm 3Dm 3p± in p-£ it m _ m _m *_ ' m 3p

3P 3Pm 3P Pm

C0SY

m

, m

x x m

where y is the angle between P and D . Ordinarily, the scan areas

'm & mm J ' are appreciably larger than the aperture area so that

9£

m

Pm

~ i

and (21b) may be written

cosy = sine cos(cf)-(() ). (22)

3D

m Similarly, it can be argued that -^— is much less than p for scan

3c|)p

areas appreciably larger than the aperture area. Thus for 6 not too

small eq. (21a) implies the critical point must be near $=<{>. Now

cosy has a minimum value greater than zero because y never reaches

'm b m

90°. The minimum value of cosy occurs at the maximum value of

'm

19
ym

= Ym *v or on tne boundary of the scan area near <J) . This maximum JU IllclA value is important because eq. (22) reveals that there are

critical points near the angle d> for 9 < 90 - v fd>1 . For > 90 - y critical points exist at max

no

cf) - <J) P

90

m m

and eq. (20) can be evaluated immediately by the method of stationary phase for double integrals [10] . Such a procedure yields after

some rearrangement

F. AD ik(d cos6+x sin9)

f- v tm m

v m

J I =

I -r^r^ e m

cos6 (23)

Actually, eq. (23) was derived using the approximation D - /(p±x ) +d

- d X

and — ^—— << 1, where x and d are defined in figure 5. The above

p dcp m

to approximations simplify the mathematics but do not alter by a great

amount the amplitude of the final expression (nor the following

conclusions)

.

The implications of eq . (23) prove to be quite significant because it shows that for 9 > 90 - v (<f>) "the magnitude of the intemax v J b

gral I is of the same order as the amplitude of the electric field

itself with each term multiplied by A D /cos9. By referring back to

eqs . (12) and (13) we see that this result implies the following:

In the re gion outside of the envel ope , which is formed by

the ra ys running from the edge of the aperture throu gh the

bounda ry of the scan area

,

the fractional error n(r) is on

the or der of unity. Thus , outside the envelope the fax

fields CO mputed from a planar : 5can in the near- field

cannot be relied upon with any con f idence

.

Moreover, diffraction from feed mounts, if present, does not affect

the above conclusion, because the radiation scattered by the feed

mounts grazes the boundary of the scan area at a wider angle than

the radiation from the edge of the aperture.

The conclusion says, essentially, that the planar scan technique does not give information about the fields outside the solid

20
angle formed by the edge of the aperture antenna and the boundary

of the scan area. (We use the term "solid angle" loosely since it

will not be a solid angle in the strict mathematical sense unless

the extension of its sides meet at a single point.) As an example

consider a circular reflector antenna of radius "a" scanned in the

near-field on a larger concentric circular area of radius a/2.

Suppose the scan area were a distance d = a in front of the aperture. Then, as shown in figure 6, the above results reveal immediately that the data from the near-field scan would not contain

reliable information about the fields outside the angle b 9 = 22.5°.

m

Newell and Crawford [11] reached the same conclusion from experimental data taken on scan planes at different distances in front

of the same microwave antenna. It appears from the above analysis

that their conclusion is a general result which holds for all electrically large aperture radiators.

It should be emphasized that the above results were derived for

the sum and difference pattern of electrically large aperture antennas and for a scan area that extends well beyond the main nearfield beam region. Also the main near-field beam has been assumed

to be characterized by planes of fairly uniform phase. The above

conclusion would not necessarily apply, for instance, to broadbeam

horns with dimensions on the order of a wavelength or less, to beams

steered nearly to the edge of the scan area and scan areas just

covering the main near-field beam, to apertures on a finite ground

plane, to defocussed antennas, or to electrically large aperture

antennas with a diverging or converging lens placed within or

directly in front of the aperture. Fortunately, special situations

and classes of antennas such as these can often be analyzed separately within the framework of the preceding analysis and results.

The special cases mentioned above are discussed in the following

paragraph.

Specifically, an analysis similar to the preceding shows that

the encircled conclusion applies to the latter three classes of

antennas (the defocussed antennas and antennas with a ground plane

or lens) , provided the scan area extends well beyond the edge of

the antenna, the ground plane or lens, and provided the edge of the ground

21
plane or lens is used as the base perimeter for the solid angle

when the edge of the ground plane or lens extends appreciably

beyond the edge of the aperture and significant scattering occurs

at these edges. Also, it can be shown [20] that the encircled conclusion applies to broadbeam horns if the center of the horn is

taken as the base of the solid angle instead of the perimeter of the

horn. (For scan areas much larger than the aperture of the horn,

there is little difference in size between these two solid angles.)

Similarly, for antennas with their mainbeam steered close to the

edge of the scan area it may be more accurate to choose a point

nearer the center of the aperture rather than the edge to determine

the side of the solid angle near the direction of the main beam.

Again it makes little difference for large scan areas. In general,

when the scan area is close to the boundary of the main beam, the

base of the solid angle outside of which the computed far-field

pattern is unreliable tends to shift from the perimeter of the

aperture (ground plane or lens) toward the center. Often the increase in solid angle is slight, however.

In brief, the above encircled conclusion (in its stated or

slightly modified forms just explained) applies to a very large

variety of antennas, including electrically large aperture antennas

operating in a sum or difference pattern (with or without beam steering, defocussing, a finite ground plane, or modifying lens) and

broadbeam horns.

Next we want to evaluate I of eq. (20) for points within the

solid angle formed by the edges of the antenna and scan area. Within

this solid angle the integrand contains no critical points of the

first kind. Consequently, the p integration can be done by parts

to yield

ik[D'-p'sin9 cos (<f>-(J> ) ]

, 2tt F (p' ,<j) )p' e m p

t-wtIJ tm p <v (24)

m o [cosy'-sinG cos(d)-cb )]

F

3D'

where again cosy' has replaced -^—^ , and the primes refer to points

on the boundary of the scan area. It should be noted that eq. (24)

represents the first term in an asymptotic expansion of eq. (20),

22
(24) remains valid for y

1 right up to 90°.

and when cosy 1 gets too small eq. (24) no longer represents a good

approximation to eq. (20). We can get an idea of the largest permissible y' by realizing that the result (24), if valid, must be much

smaller in amplitude than the amplitude of the integrand of eq. (20)

multiplied by the change in distance p as D changes from D' to

D' + A/2. A little mathematics shows that this condition is always

m

y~Y

7 satisfied if cosy' is greater than about V-p- , where d' is the perpendicular distance from the edge of the aperture to the scan plane.

However, when used to find an upper bound expression for |l| (see

eq. (28) below) , eq

Eq . (24) has been derived under the condition that the fields

emanate or at least appear to emanate from points on the edge of the

antenna aperture. If the fields are also scattered from the feed

mounts, eq. (24) must include these fields as well

associated with these mount scattered fields will always be equal

to or greater than that of the edge diffracted fields--since the

fields scattered by feed mounts make larger angles with the plane

of the scan area at its boundary than the edge diffracted fields.

Thus, the integration by parts remains valid (as an upper bound expression within the solid angle formed by the edges of the aperture

and scan area), and eq . (24) holds even when there exists appreciable

radiation scattered from feed mounts in front of the antenna.

For angles near the z-axis (sin0 << cosy ) eq . (24) becomes

The cosy 1 'm

I =

\

2tt

7T1 J

F + e tm

ikD'

m

cosy'

'm J

p' e ikp' sin9 cos ((f) -(j) ) d<J> (25)

Upon taking the amplitude of eq. (25) we find

, 2tt ih ! 2tt

F. e tm

ikD'

m

cosy

m

p' d(f) . P

(26)

For scan areas whose boundary is well outside the edge of the aperture, cosy' - 1. For scan areas with boundaries somewhat close to

' 'm

the edge of the aperture, the term (in the summation over m) which

corresponds f to the minimum D' (maximum y') predominates (assuming

m

v. i m

; f to

23
fairly uniform illumination around the edge of the aperture) , so that

Im

F + e tm

ikD'

m

cosy'

'm cosy m

'max

^ F

t

ikD'

m

m

In either case, we see from eq . (19) that

tm iyp',y

ikD'

m

cosy'm cosy

max

where F (p',4> ) is the electric field at the boundary of the scan

area Eq . (26) can now be written

it-i . A

2tt cosy

max

2tt Jo E

t

(p',<f> )|p' d(j) , (27)

or

A L

max

2 cosy

max

(28)

max

with |E c denoting the maximum amplitude of the transverse electrie field found on the boundary C of the scan area, and L is

the maximum width of the scan area. If the output of the measuring

probe at the limits of the scan area is down at least X dB from its

maximum output, then eq. (28) may be rewritten

A L

max 10

X_

20

2 cosy

'to' (29)

max

where E represents the highest amplitude of the transverse electric

tO

y g field found on the scan area. ' Equation (29) combines with eqs

.

(13) and (14) to yield an upper bound expression for the error n in

'For an arbitrary probe (i.e., not necessarily a dipole probe) E-to

represents the highest output amplitude of the probe on the scan area

and X the largest output amplitude of the probe at the edge of the

scan area measured in dB down.

^The maximum electric field on the scan area in the near-field

(z << A/A) of an electrically large aperture is very nearly equal to

the maximum electric field on the aperture itself.

24
the far-field produced by neglecting the fields outside the finite

scan area:

__X

, T max ., n 2 „-f=\

nCF)<-A-L 10 _gW 9 (30a)

E

t

2 cosy If ^— da I

1 max ' i E .

' A to

o where g(r) is the ratio of the amplitude of the maximum far-electric

field to the far-electric-field in the given direction r. In other

words, it is the inverse of the normalized far-field pattern. Use has

been made of eq. (10) which shows that for sum patterns

Aril" I - 1/ E\ da I , (30b)

o where |E I is the amplitude of the maximum far-field, and the

o r

-*"00 integration is performed in the near-field over the area A , that

part of the beam which has nearly uniform phase. Since we are in

the near- field, A - A cos0 , where A is the aperture area. (The

factor cos0 accounts for the reduction in effective aperture area

o

r for beams which are steered off-axis electronically through an

angle .

)

6 o Because |E. 1 I/E < 1, where E is the highest amplitude of elect ' o — ' o

6 r trie field on the scan area, and E.too- E cos0 , we can write

o

= a cos0 /A = a/A, (31)

J E^- da

A

c to

o where the factor a is greater than its minimum value of 1.0 (for

apertures of uniform amplitude and phase) but less than 5 for most

tapered aperture distributions found in practice. For example, the

tapered distributions found in Table IX of [13] have a maximum a of

4.0. Newell [17] has found that the factor we have called a has not

been greater than about 2 A/A for any microwave antenna he has

measured. Since the effective area A (see [12] for a definition

of effective area) is less than the aperture area A and greater

than . 5A for most aperture antennas [13] , the experience of Newell

25
also indicates that a is less than 4 or 5 for nearly all electrically

large aperture antennas found in practice.

Equation (31) combines with eq. (30a) to give the final expression for n(r)

nO)

__x

, T max t A 2 r —-, a A L 10 g(r)

2 A cosy

max

(32)

max

max

where

A = area of the antenna aperture.

A = wavelength.

= maximum width of the scan area.

= maximum acute angle between the plane of the scan area and

any line connecting the edges of the aperture and scan area.

= the largest amplitude of the probe output at the edge of the

scan area, measured in dB down from the maximum amplitude of

probe output in the scan plane

.

= a "taper" factor--equal to a minimum of 1.0 (for apertures of

uniform amplitude and phase) and less than 5 for most tapered

distributions found in practice. (See eq. (31) for the precise definition of a.)

g(r) = ratio of the amplitude of the maximum far-electric-field to

the far-electric-field at the given direction r, i.e., the

inverse of the normalized far-field pattern. (g(r) = 1 for

the center of the main beam, or beams if a difference pattern.)

(If desired the errors in the gain function sidelobe level, polarization ratio, and beamwidth may be calculated from eqs . (2) -(5) once n and the far- field pattern is known .

)

Equation (32) has been derived for antennas operating in a sum

pattern. But since a difference pattern can be divided into two,

approximately equal, sum patterns with wavefronts slightly skewed,

eq. (32) holds for difference patterns as well. (For a difference

pattern one should still use the taper factor a of the constituent

sum patterns .

)

In summary, eq. (32) applies to either sum or difference patterns of all electrically large aperture antennas (including antennas

26
with their boresight direction steered away from the axis perpendicular to the scan area) within a solid angle (sine << cosy ) about

the axis perpendicular to the scan area. Again, it has been assumed

that the scan area extends well beyond a main near-field beam which

is characterized by planes of fairly uniform phase. It can be shown

that eq. (32) and eq. (36) below apply even to defocussed antennas,

apertures in a finite ground plane, and to antennas with a diverging

or converging lens, provided the angle y is chosen in accord with

max

the second paragraph preceding that of eq . (24). The condition

(sine v << cosy ) can be made more specific by returning to eqs. max ^ J t>n

(24) and (25). For all practical purposes, eq . (25) follows from

eq. (24) if

sine < ~- cosy (33) = T sinf

2 ' max 2 max

(6v = 90 - y ) max 'max

For example, if y were 45°, condition (33) becomes 6 < 20°, which

is a large enough angle to include many side lobes of most microwave

antennas (assuming their boresight direction at = 0) . Roughly

speaking, eq. (32) represents a valid upper bound within the region

< i

2 max

As an upper bound, eq. (32) remains valid for y right up to

max

90°. However, from the discussion immediately following eq. (24),

it is unlikely that eq. (32) would remain small enough to be very

useful when

(34)

where d refers to the minimum perpendicular distance from the mm r r edge of the aperture to the scan plane.

The error n given by eq . (32) can be compared with the results

of the empirical error analysis performed by Newell and Crawford [11]

They took "centerline" data on a near-field scan plane 25 cm in front

of a circular, fixed-beam "constrained lens" array, which was 80 cm

in diameter (see figure 9). The centerline was 213 cm long.

Assuming a rectangularly separable field pattern, they first

used the 213 cm centerline data to compute the far-field pattern.

Successively, more and more of the 213 cm distance was deleted and

27
the corresponding far-field pattern computed. In that way they

could get an idea of the errors involved in scanning on a near-field

plane of finite area.

The envelope of their near-field centerline amplitude is reproduced in figure 7a. The on-axis gain change computed by Newell and

Crawford by deleting distances from the scan line is reproduced by

the dashed line in figure 8a. The solid lines in figure 8a represent

the maximum envelope of on-axis gain change calculated from eqs . (32)

and (2c) of the present report. The values of X and L

max (see eq.

(32)) were taken from figure 7a. The value of a was estimated from

eq. (31) and figure 7a to be about 3. The remaining parameters

needed to calculate n from eq. (32) are contained in reference [11]

:

* -if d

Y „ = tan t r max

T max

—* a U z

J J

A = 3.26 cm

A = ira 2

a = 40 cm

d = 25 cm

g(r) 1.

Figure 8a confirms the result that the fractional error r\ of

eq. (32) represents a reasonable upper bound. In the region

max

L /2a > 1.7 (y < 42°) the upper-bound error from eq . (32) is

max

no more than double the error estimated by computer "deconvolut ion"

of the near-field data. The upper bound error grows inordinately

mo y large, however, for L

' /2a much less than about 1.2 (y > 75°),

max

as eq. (34) predicts.

Figure 8b shows the same comparison as that made in figure 8a

but for a 46 cm (18 in) reflector antenna operating at 60 GHz (A =

.5 cm). (The envelope of the amplitude for a centerline scan of this

antenna is shown in figure 7b. The value of a is about 2.) The

dashed line in figure 8b represents the on-axis gain change which

Newell [17] computed by deleting distances from the centerline scan

length taken 43.18 cm in front of the aperture. The agreement between his computations and the upper bound solid curve calculated

from eq. (32) is even closer than for that of the constrained lens

28

H
antenna (figure 8a). The upper bound error is no more than double

the computed error for L /2a > 1.05 (y ov < « ) which is very

max

close to the value of y1 < 84° predicted for the range of usefulmax r &

ness by eq. (34)

.

It appears from this somewhat limited experimental evidence with

centerline data that the simple formula (32) provide s a useful upper

bound error at least for scan areas with cosy > 'max

It should

be noted, however, that centerline data do not accountror changes in

phase of the field around the perimeter of the scan area, and thus

would predict larger finite scan errors in most cases than the complete 2-dimensional scan data. An upper bound expression of smaller

magnitude generally than eq. (32) that takes the phase changes into

account can be derived by returning to eq. (24). Under the condition

of eq. (33), eq. (24) becomes

— 2tt cosymax

2tt / E

t

(p',<j) )e

o ^

-ikp'sin9 cos (((>-<}> ) p 'p' d<|> (35)

Again E (p',(J> ) refers to the transverse electric field, i.e., output

of the dipole probe, at the boundary of the scan area. Substitution

of eq. (35) into eq. (13) yields the fractional error n (r) in the

far-field for 1 max

(90 Y ) 'max; '

nO) ;

2tt -ikp'sinQ cos(<J>-cf) ) J E

t

(p',cf> )e P p» d*

27rr I

i E (r) ^ j

I

i cosy

r->oo 'max

(36)

The far-field pattern r|E(r)| in eq. (36) can be approximated

AE

to

r ^°° analytically by — or found by deconvoluting the measured nearaAg(r) _

field data. The field E t

(p',(j> ) at the boundary of the scan area

z p

can be taken from the measured near-field data. Thus, in practice

both the numerator and denominator of eq. (36) can be determined

straightforwardly. Although eq. (36) involves more computations than

eq. (32) even for the on-axis value for which 6=0, it could be computed by a simple routine added to the program which deconvolutes the

near-field data, since for arbitrary probes E (p',cj) ) can be

replaced by the output of the probe (for two orientations, in

29
general) en the perimeter of the scan area. In cases where the antenna

pattern is assumed separable in xy coordinates and only centerline

data is taken, eq. (36) cannot be applied but eq . (32) still can.

9

In Table 1 are listed the errors calculated from eq. (32) for

some of the far-field parameters of a typical X-band antenna operating in a sum and difference pattern. The finite scan errors are

proportional to wavelength, so changing the wavelength while holding

the other antenna dimension the same merely changes the values in

Table 2 proportionately. Of course, such an isolated change is

rather unrealistic.

B . Position and Instrumentation Errors

To determine the radiating fields of an unknown antenna by

scanning on a near-field plane with a given probe, the output b ' (P)

of the probe must be recorded throughout the scan area A' (see footnote 3a). In principle, the data points should lie in a plane and

the position of the probe should be recorded exactly as it scans

from point to point. And, ideally, the instrumentation used to

measure the phase and amplitude of b' should do so with perfect

accuracy

.

Obviously, in practice, neither the position of the probe nor

the phase and amplitude of the probe output b' can be measured

exactly. Regardless of how small the uncertainties in the measurement of b'(P) they will introduce errors into the calculated farfield. It is the purpose of this section to derive general expressions which estimate the magnitude of the errors in the far-field

produced by the inaccuracies in measuring the position and output

of the probe in the near-field.

Specifically, we want to evaluate AE in the far-field so that

the fractional error n(r) of eq. (1) can be determined. (In Section

y It is interesting to note that ri in both eqs . (32) and (36) does not

approach zero but simply an insignificantly small number as the scan

boundary approaches infinity. This limiting value of r\ represents

the contribution of the oscillatory part of the Fourier transform mentioned in footnote 3b. Consequently, once the edges of the scan area

reach the region where the fields behave as 1/p', there may be no

advantage to scanning on a larger area at least if only the pattern

near the boresight direction is required.

It is shown in reference [20] that eqs. (32) and (36) slightly

modified apply to broadbeam horn antennas as well as electrically

large aperture antennas. For these broadbeam antennas the limiting

value of n can be significant.

ilk
II the errors in gain function, sidelobe level, polarization rati*,

and beamwidth are derived in terms of n(r) and the far-field pattern.) In order to simplify the theoretical analysis, the errors

will be evaluated as if the probe were a perfect electric dipole.

The justification for choosing a perfect dipole in the analysis is

similar to that stated in Section III. A. The dipole measures the

electric field components at a point. All physically realizable

probes respond to a weighted average of the fields near the probe.

Thus any small error in position would be expected to change the

output of a perfect dipole by as much or more than any other probe.

As for instrumentation errors, they remain essentially independent

of the particular measurement probe. Also, as in the previous sections, any uncertainty in probe receiving characteristics (S^-, ) will

be ignored for this part of the error analysis.

Under the above conditions the far-field error in AE may be

found with the help of eq. (10):

A E .(F) = - ik C0S6

e ik(r " d C0SG)

/ AE t (P,d)e~

lFR ' P

dP. (37)

Z

2ttt A'

r Y -> oo

AE (P,d) is the difference between the actual electric field at the

point (P,d) and the measured output of the hypothetical dipole probe

(for two orientations in general) at the point (P,d). For errors near

rn O -y" the z-axis (0 < 2A/L , see footnote 5) eq . (37) becomes

iic iicrr <n - -

"i^R-P

AE(r) = -i£- e 1Kir " aj

J* AE (P,d) e r dP, (38)

2i\r A

r -* °° o where A designates that part of the scan area over which the major

variations in phase are relatively small. (For the position and

instrumentation error analysis the scan plane is assumed to lie

nearly parallel to the near-field planes of "uniform" phase, i.e.,

perpendicular to the center of the main beam of sum patterns or to

the null axis of difference patterns.) For scan planes in the very

near-field, A is approximately equal to A cosG , i.e., the projected

area of the antenna aperture. (Only electrically large aperture

31
antennas are being considered.) 6 is the angle between the perpenO /\ dicular to the aperture and the perpendicular (e ) to the scan plane.

0=0 for beams which are not steered off-axis electronically.

The scan area outside A can be neglected for this part of the

o error analysis because the rapidly changing phase in the region outside A contributes little when integrated to find the far-field and

the errors in the far-field near the z-axis. It follows from footnote 5, or more rigorously from an asymptotic analysis like the one

performed in Appendix A for a circular aperture of uniform distribution, that eq. (38) can be used as an upper bound expression for AE

in the region given approximately by < 2A/L where L is the

maximum breadth of the partial scan area A . This region is large

enough to include the first sidelobe maximum of many electrically

large aperture radiators. For example, a circular aperture of uniform

distribution and radius a - L /2 has its first sidelobe maximum at

o an angle - 1.7X/L radians.

Physically this 2A/Lmax condition says that for most electrically

large aperture antennas, the part of the near-field (A ) over which

the phase is fairly uniform strongly influences the far-field within

the angle 2A/Lmax

. Beyond this angle the edge diffracted fields

dominate the far-fields to a greater and greater extent until in the

far sidelobes the fields are determined essentially by the edge diffracted fields alone.

To find either the position or instrumentation errors, the integral

-i^R-P

T =

J AE.(P,d) e r dP (39)

A

z o

in eq. (38) must be evaluated. Of course, the integration can be

performed only after AE (P,d) is found. In Section 1 below, AE

t

, T , and thus n (r ) are evaluated for position errors, and in Section

2 for instrumentation errors.

The approach that is taken is quite straightforward. For position errors AE is expanded in a Taylor series about (P,d) assuming

the deviation in the position of the scanner is small compared to a wavelength. The Taylor series and the integral (39) into which it

32
is substituted divides naturally into a longitudinal or z-position

part and a transverse or xy-position part. The upper-bound for

each integral and thus for the z- and xy-position errors are then

determined as a function of the measured near-field data and the

computed far-field pattern.

For instrumentation errors, AFT. is expressed in terms of the

amplitude and phase errors introduced by the nonlinearities in the

receivers which measure these quantities as the probe traverses the

scan area. Mathematically, the integrals involving the receiver

phase errors are handled in the same way as z-position errors, and

thus the final upper-bound expressions have the same form. The

integrals involving the amplitude errors have their upper-bound

determined by characterizing the receiver nonlinearity in measuring

amplitude in dB of error per dB down from the maximum amplitude of

the probe output on the scan area.

1. Position Errors

Consider the scanner which moves the probe throughout the nearfield scan area. Typically the scanner covers the area by traversing a grid or raster of lines while the probe output is recorded at

given points along each line. Ideally, all the scan lines lie perfectly straight and parallel in a single plane, and the position of

the data points along each line is recorded exactly. In reality,

of course, none of these idealizations hold, basically because the

scan lines will never be perfectly straight and the data must always

be recorded over an interval rather than at a point.

Regardless of the reason for the position errors, they all

effect the near-field data simply by positioning the probe at points

(P+AP, d+Az) rather than (P,d) . In other words, the difference

AE (P,d) in eq. (39) can be written for position errors as

AE

t

(P,d) = E

t

(P+AP,d+Az) - E

t (P,d), (40)

where, as usual, E (P~,z) is the electric field at the point (P,z) in

the near-field. In general, AP~ and Az, which will be referred to as

displacement errors, are functions of the transverse position P.

33
Before continuing with the analysis it should be pointed out

that displacement errors caused by a small initial translation of

the entire scanner with respect to the test antenna will not cause

a change in the computed far-field amplitude, as eqs . (6) and (8)

indicate. Also, an initial rotation of the entire scanner through

a small angle will have no effect on the far-field pattern other

than to rotate the entire pattern through that same angle.

Since the displacement errors must be much smaller in magnitude

than a wavelength, the right hand side of eq . (40) can be expanded

in a three dimensional Taylor series. By letting Ar = AP+Aze and

keeping only the first two terms in the series, eq. (40) becomes

AE

t

(P,d) = Ar-VE

t

(P,d) + j Ar • [Ar «VVE (P,d)

]

(41)

where VVE^ = (WE )e + (WE )e

t

v x^ x

v y y

and VE\ = (VE )e + (VE )e . t

v x

J x y y

Substitution of eq. (41) into eq. (39) yields a useful expression

for I ,

1 -i-R-P

= / Ar- [VE. +yAr-WE. ] e r dP.

A

z L z o

(42)

Again it is emphasized that Ar is, in general, a function of P, the

transverse coordinates over which eq . (42) is integrated. Also, as we shall see shortly, it is necessary to retain the second term in

the integrand of eq. (42) when on-axis errors for sum patterns are

considered. As a check on eq. (42) let R = (on-axis) and Ar be

constant over A so that Ar can be taken outside the integral. The

o terms containing the transverse part of the gradient operator convert to line integrals around the boundary of A . These line integrals must equal zero because in effect eq. (42) assumes negligible

fields outside A . Only the "z" derivatives, which can also be

taken outside the integral, are left, and eq. (42) may be written

34

^B
Az |- + Az 2 |-r 8z 3z / E dP.

A

z o With the aid of eq. (10) , eq. (43) converts to

(43)

I = -(Az+ikAz 2 )2^re lk ( r

~ d ) E^ (r)

o

J t

K J

X" -> oo

(44)

Equations (38), (39), (42) and (44) combine to show that

E(r)+AE(r)

T-X30 1 + C^Az)

8

E(r)

f -»-00 *

i.e., the error in the far field amplitude calculated from eq. (42)

I A -p I is of higher order than (-U-

<-) 2

, i.e., it is negligible when Ar is

constant -- a result which must hold if eq. (42) is a valid expression for I , because, as mentioned above, a translation of the entire

scanner has a negligible effect on the far-field pattern. Equation

(42) also checks in a similar way for R f 0, but the proof is more

involved.

In order to evaluate T of eq. (42) exactly, both Ar and E

T

(F)

in the near- field would have to be known. Fortunately, an upper

bound approximation can be found for T without detailed information

about Ar or E (r)

.

Consider one component, say E , of the integrand of eq . (42).

For an electrically large aperture antenna, we can express E within

.A. the area A as

o

E (P,z) * (1+AA (P,z))E (P) e i(kz + cJ) ox

+Act> x (P,z))

(45)

where d>r is a real arbitrary phase constant, and AA , Ad> and E

ox J r x' r x ox

are real functions of the indicated near-field coordinates. The

functions AA (P,z) and A<J) (P) typically oscillate over a distance

equal to or greater than a wavelength, each with a magnitude usually

much less than one. E (P) is the smoothly tapered amplitude function. In other words, eq. (45) simply states that the near-field

across the area A of an electrically large aperture antenna has a phase equal essentially to kz and a smoothly tapered amplitude except

for small variations which oscillate over distances equal to or

35
greater than a wavelength. (Recall that for the position and instrumentation error analysis we are assuming that the perpendicular to

the scan area is approximately aligned with the boresight direction

of the antenna.)

In Appendix A eq. (45) is verified for a circular aperture of

radius "a" and uniform aperture distribution (E = E ) . In that

1 ^—r— ox

case AA and A<|> are on the order of — /A/a or less. Rusch and

Potter report similar results for the circular aperture [14] . Equation (45) has been verified experimentally by the substantial nearfield scan and extrapolation data taken at the NBS . The measured

amplitude and phase on a near- field plane of a typical microwave

antenna are plotted in figures 9a,b,d. This particular scan was

taken 25 cm in front of the circular "constrained lens array" (see

figure 9c) of radius 40 cm operating at 9.2 GHz (A = 3.26 cm). This

antenna is the same one described in the paragraph following eq. (34)

and whose centerline amplitude envelope is plotted in figure 7a.

When eq. (45) is substituted into the integrand of eq . (42) and

all terms higher than either second order in |Ar|/A or first order

in |Ar|//A~ are discarded, the following expression for the x component of I remains

I - ox */ [Ar-V|E

x

2ttAz

X ox

+ i

'2ttAz'

A

I j .

i*. -i-R-P

r

where E

i4>.

dP,

(46)

x

E e x

or from eq . (45)

|E I = fl + AA )E 1 X ' v x

; ox

d> = kd + y <J> + A* . x

Y ox r x

It has been assumed that the transverse (P) and longitudinal (z)

displacement errors are of the same order of magnitude. Eq. (46) can

i<j> i(kd + (j) ox ) be simplified further by writing e as e (cosAcfj^+i sinA<|> x

) , to put r I in the form

ox

36
i(kd+<f) ) I * e OX r ( Ar . v | E

I _ 1

5

2 E + i6E -j ox i K ' x ' 2 ox ox ; o -i^R-P

• (cos A(f> + i sin A<|> )e

r dP

.

(47)

(6 = 2-ttAz/A)

First consider on-axis errors for antennas operating in a sum

mode, that is ^-- << 1 or 6 less than about A/ (10 L

max

) . Then the

exponential in eq. (47) can be approximated by unity. The arbitrary

phase r constant i> can be chosen so that Ad> varies about zero

ox T x throughout the area A . Furthermore, since the variations in Ad> te o

T x are small, oscillate many times across the area A , and remain completely independent of the variations in the displacement errors Ar

of the scanner, to a high probability, c|> can be chosen such that

the integration in eq. (47) which is multiplied by sinAcj) can be

neglected compared to the maximum possible value of the cosAcf> integration. In addition, since E cosAc}> remains positive throughout

A for a sum pattern, the reference plane (z=d) for 6 can be chosen

to make

J 6E cosAcJ) dP = 0. (48)

; ox Yx *• J o Thus, under the above conditions the iS term of eq . (47) is eliminated entirely and eq. (47) simplifies to

i(kd + <f> )

n I = e ox

j ( Ap . v E

_ i

6

2 E ^ cos A(J) dp> ox i

v t ox 2 ox ; x

o

(A*• = A +e |-)

t Z dZ J

(49)

E in the first term of eq . (49) has replaced |E I =(1+AA ) E of

ox n v J r ' x

1 v x^ ox

eq. (47) because the many oscillations of AA would also (to a high

probability) eliminate upon integration all but higher order contributions from the V(AA v ) term.

x'

It is interesting that eq. (48) expresses the same condition

chosen by Ruze [15] in his well-known work on "antenna tolerance

theory," and under the above assumptions the z-position errors

always lower the on-axis gain value of the sum pattern.

Of course, in his work 6 represented a small "arbitrary

37
phase error or aberration" (what we call A<J> ) rather than the displacement errors of a near- field scanner. It would be desirable that

the reference planes of Ax and Ay (AP = Axe +Aye ) be chosen such

that / AP«V E cos A<|> dP" became zero also. Unfortunately, such a A

Z 0X X

9E

choice is impossible (in general) because neither derivative, ox

8E 9x

or

ox

, stay the same sign throughout A . From eqs . (38) and (39) it is seen that the fractional error n of eq. (1) can be written as

n(r)

Ar |E(r) ox

+ |I | 2

1 oy

X*->oo

(50)

The amplitude of I is found from eq . (49) to be

|I | = |/ (AP-V.E - \ 6

2

1 E )cos Ac}) dP OX ' A t OX 2 OX ; X

o

<_ AP

X6

/ |V^E I dP + \ 6

2 max i ' t ox ' 2 max i ox A

/ E_ dP,

A

o

(51)

where AP and

max 2ttmax

are the maximum magnitudes of the transverse

(P) and longitudinal (z) displacement errors, respectively, introduced by the near- field scanner.

The last integral in eq. (51) can be related to the maximum

far-field |E I , i.e. from eq . (10) or (30b)

xo r-*-°o 7

/ E dP = ArlE I ^ ox ' xo ' r^-°° o

(52)

The remaining integral,

JA

V^E dP,

t ox ' ' (53)

is a bit more troublesome. However, it can also be estimated by

expressing the integral in polar coordinates (p,<J>) as follows:

/ |V.E I dP = f A

t ° X A

o o

3E

ox

3P

-, 8E

1 ox

^ 2

P 94)

pdpdcj)

(54)

38
If we assume that the amplitude of the aperture distribution tapers

with much greater slope in the radial direction (p) than in the azimuthal direction (<f>) , the second term under the radical sign of eq

.

(54) can be neglected, leaving

9E

_ f ox

A

o 3P

pdpdcj)

.

(55a)

9E

The negative sign is present in eq . (55a) because ox

3p

is negative

for amplitude distributions which taper toward the boundary of A . (We are assuming p = at the maximum of E .) Integration of eq.

O -A. (55a) by parts with respect to p yields

8E

•JA

ox

3p

where E and L

xo o

omax

pdpdcj) =

J E dpdc}>

A

ox

o

tt_ r T max 2-C 1j * xo o (55b)

denote the maximum amplitude of E and the maxiox

mum breadth of the partial scan area A , respectively. With the aid

of eq. (55b), eq . (54) becomes

/ |V F | dP < £ L

max

E

{ ' t ox 1 — 2 o xo

o

(56)

Since the area A is in the very near- field of the aperture antenna,

the maximum amplitude E on A is approximately equal to the maximum amplitude on the aperture itself.

maximum far-field by (see eqs . (30b) and (31))

Thus E is related to the

xo

aAr E

xo ' r- xo A

and eq. (56) becomes

(57)

J |V^E I dP <

{

' t ox 1 —

o

, T max i ~ i TTaArL E

o ' xo ' r-»°°

2 A

(58)

Substitution of eqs. (52) and (58) into eq. (51) yields

r . „ . max

TraAP n L I I < ArlE I OX ' — ' xo

'

]f->OQ

TTaAP 1/

max o

2 A

+ i 6

2 2 max (59)

39
A corresponding r inequality holds for ll I with IE re-

? n 7 ' oy '

' yo ' r->°° placing |E | ^oo

. At first sight, one may raise the objection that

the reference plane for <5 cannot necessarily be chosen so that both

eq. (48) and the corresponding equation with E are satisfied

simultaneously. It should be noted, however, that the two equations

need not be satisfied simultaneously. It is only necessary that one

reference plane can be found that allows eq . (48) to hold, and that

a second reference plane can be found that allows the corresponding

equation with E to hold. If the two reference planes are a slight

distance apart, the relative phase of the x and y components of the

far-electric-field will be in slight error but not the Hermitian

amplitude of the far-electric-field, i.e., the far-field pattern

will remain unchanged even though the far-field polarization ratio

will be shifted slightly.

Eq. (59) and the corresponding equation for |l | combine with

eq. (50) to give an expression for the maximum position error n in

the far-field:

n(r)

An T max

TraAP L -. max o 1 x-2 + -7T 2 A

2 maX

A

J 10Lmax

o

(sum patterns)

(60a)

Equation (60a) was derived assuming the antenna was operating

in a sum pattern and does not apply to difference patterns since eq.

(48) may not be satisfiable for difference patterns. However, in

Appendix B it is shown that an error expression similar to eq . (60a)

may be derived near the boresight direction (null axis) of difference

patterns as well. Specifically,

n(r)

An T max

TraAP L

max o

2 A

+ 4AF6

max

g(r), 6 <

J 10Lmax

where AF and g(r) are defined below after eq. (61)

Finally for the angular region > A/10 L

max

o

(difference

pattern)

(60b)

but still less

than 2A/L (see eqs . (37) and (38)), we can derive an upper bound

expression similar to eq. (60a). Specifically, the magnitude of I Qx

in eq. (47) may be written

40
J<| |Ar-V|E || dP + \ j |6E | dP.

A

X A

0X

o o ox

The term second order in <5 has been ignored since now the first order

term in 6 is the major contributor to the z-displacement error. When

we carry the analysis through in a way similar to that described

between eqs . (51) to (59), the following expression for the fractional error for both sum and difference patterns results :

n(r) <

An T max ^ TTO.AP L

max o o g(r)

2

A

» max 1QLmax

2A

max

o

(60c)

Although eq. (60c) was derived for 9 < 2A/L , it remains valid as an upper-bound expression all the way to = tt/2. This result

follows from the fact, which will not be proven here, that the contribution from the integration in eq. (37) outside the area A is

negligible compared to the upper-bound contribution obtained from

inside A and expressed by eq. (60c). (It is interesting to note

that the maximum possible errors in even the far sidelobes are

determined by the displacement errors across A , whereas the field

itself in the far sidelobe region is determined by the near-field

outside A .) o

' Thus, if we combine eqs. (60a), (60b), and (60c); insert the

approximations, L

A

2ttAPmax

max

o

I

max A (£

max

) 2 into eqs. (60); and let

max A far-field hemisphere emerges

an upper-bound expression for r\ valid over the entire

n(r) < A + n max max l z 2£

6

2max

g(r)

2

^z =( 8AF 5 max

max

(sum patterns)

(difference patterns)

(sum and difference

patterns)

10£ max

10£max

<e<

(61)

41
where

X = wavelength

„max = maximum width of the antenna aperture

ATMa Y

= 2lTAPrr1 o V / A » where AP is the maximum amplitude of the trans- IlLdA IlLdA IIldA verse displacement errors within the partial scan area A . o (A is that part of the scan area over which the phase is

fairly uniform. For near-field scans parallel to the aperture A - A, the aperture area.)

6 m o V

= Z^^z /\ , where Az m „ v is the maximum amplitude of the IIldA IlLdA ITldX longitudinal displacement errors within the partial scan

area A . o AF = fractional difference between the amplitude of the two main

far-field lobes of the difference pattern (see Appendix B)

a = a "taper" factor -- equal to a minimum of 1.0 (for apertures

of uniform amplitude and phase) and less than 5 for most

tapered distributions found in practice. (See eq . (31) for

the precise definition of a; for a difference pattern one

should still use the taper factor of the constituent sum

patterns .

)

g(r) = ratio of the amplitude of the maximum far-electric-field to

the far- electric- field at the given direction r, i.e., the

inverse of the normalized far-field pattern. (g(r) = 1 for

the center of the main beam, or beams if a difference pattern.)

(If desired, the errors in the gain function, sidelobe level, polarization ratio, and beamwidth may be calculated from eqs . (2) -(5) once

n and the far-field pattern are known.)

The expression (61) represents an upper bound to the far-field

error n caused by inaccuracies in the position of the near-field

scanner. It applies to both sum and difference patterns in the

entire forward hemisphere of all electrically large aperture

antennas. Equation (61) was derived under the assumption that the

scan plane is parallel or nearly parallel to the near-field planes

of "uniform" phase, i.e., the plane perpendicular to the electrical

boresight direction.

In addition, eq. (61) holds for arbitrary (random as well as

systematic) errors in the positioning of the scanner, since only the

42
maximum magnitude to of displacement r errors A and 6 are required max max l to evaluate eq. (61). However, it proves useful to derive an expression similar to eq. (61) which separates the effects of the

systematic and random errors in position.

To do this and also clarify what is meant by systematic and

random errors, consider the motion of the scanner as it takes measurements along lines in the near-field plane. As the scanner moves

along each line it will deviate from a perfectly straight -line by a gently varying curve that will contain at most a few oscillations

from one end of the scan line to the other. These curves, which may

also change gently from scan line to scan line, represent the systematic errors in position of the scanner. For example, a slight

warp or deformation of the scanner frame would create a systematic

error

.

Superimposed upon the systematic deviations from the straightline would be position errors which changed randomly (within limits)

from measurement point to measurement point. These random errors in

position have zero or nearly zero mean and could result, for example,

from vibrations of the entire scanner or from a slight play in the

drive mechanisms. The maximum magnitude of the random errors are

often smaller than that of the systematic errors. However, it is

possible that the scanner is aligned so precisely, that essentially

all but the random errors are eliminated.

Return to eq. (47) and separate AP and 6 into systematic and

random displacement errors, i.e.,

AP = AP S + AP

rn (63a)

5 = 5 S + 6

rn

. ( 63b ) —rn All the integrals in eq. (47) which contain linear terms in AP or

t*ji rn rn

<5 can be dropped. Since AP "" and 6 change randomly from measurement point to measurement point (over distances less than A/2) , the

integrals actually summations -- see footnote 3a) containing these

linear terms will be extremely small compared to the largest possible

errors produced by the remaining integrals (summations). Thus, substituting eqs . (63) into eq. (47) and proceeding as we did before

with eq. (47), yields the desired expression for n separated into

43
systematic and random errors,

n(r)

aX . s 21max max

s ,rn g(r)

2

(6 ) 2 +(6 in

) 2 (sum patterns)

max' v max' r '

i S ' rn

=< 8AF 6

:max

max

(difference patterns)

(sum and difference

patterns)

6 <

X 1(Umax

A — <e<

10JI

max

(64)

All symbols have the same definitions found after eq. (61). The

superscripts "s" and "rn" refer to "systematic" and "random" errors

respectively

Note from eq. (64) that to the given order of approximation the

longitudinal random errors (6 ) , but not the transverse random

errors (A ), cause an error in the on-axis far-field of sum patr n terns (provided, of course, that A is of the same order of magnirn tude or less than 6 ). In addition, eqs . (64) and (61) above show

that the maximum possible transverse (xy) position errors do not

depend upon wavelength for a given g(r) since A behaves as 1/X.

max

Also note that in expressions (64) and (61) above, the z-position

ma v error is not continuous across the angle 6 = X/lOi . There lies

no contradiction in this fact since the expressions remain valid as inequalities. The jump between the two regions merely indicates

that the upper-bound was determined by a different method in each

region. Obviously expressions (61) and (64) do not represent a

rn o -y- least upper- bound throughout the region 9 > X/KU . This region

will be discussed further at the end of the section.

One can get an idea of the magnitude of the position errors by

plotting the on-axis gain from eqs. (64) and (2c) for the fixed-beam

constrained lens array described in the paragraph following eq. (34)

and shown in figure 9. In that case,

A/£max = .04

g(r) = 1,

44
and with a equal to 3, eq. (64) combines with eq. (2c) to give

s nf < ± 8.7 A" + J-(^S

) max 2

K max ; 2^ max ; (65)

I£ we assume 6

written

rn

max

= and let 6=6'

o max

= A"max

eq. (65) may be

dB

< ±i 7 6 o (.03 + f

« ).* (66)

dB

particular r antenna, 6 must be less than about ' o

The solid curve in figure 10 depicts the maximum value of r\ r in

u eq. (66) versus 6 , i.e. when the random errors are negligible. The

o dB s s dashed line represents the maximum error r\ n when the A = 6 = r G max max

r n and 6=6 , i.e., when the systematic errors are negligible,

o max' ' J & &

To insure that the on-axis gain is less than .01 dB for this

027. That is, each

component of systematic displacement error of the scanner must be

less than .027 A/2tt, or less than about .004A. For this constrained

lens antenna A = 3.26 cm, so the errors in each component of displacement must be less than about .14 mm for better than .01 dB

accuracy in the on-axis far-field.

For random errors only (the dashed line) the z -displacement

errors must be less than about .26 mm for .01 dB accuracy in the

on-axis far-field for this particular wavelength. As a matter of

fact, when the systematic errors are negligible, eq. (64) shows that

dB the "random" error in gain & x\ n for the sum-pattern main-beam of an G,rn r arbitrary (electrically large aperture) antenna can be written

simply as

dB < + 8^ f6 rn . 2 * G,rn — 2 ^ max-'

(6

rn

= 2TiAz

rn /A) *• max max }

(67)

dB rn rn For x\ n < ± .01, 6 must be less than about .05 radians or Az G,rn — ' max max

less than .008 of a wavelength (3 degrees). That is, the random

errors in z-position should be no greater than about ± .01A to insure

a .01 dB accuracy in the gain of the main beam. Equation (67) reveals that the random error in the on-axis far-field gain of sum

patterns increases as the square of the random error in the

*Actually, as mentioned above, the z-position or phase errors always

lower (under the stated assumptions) the on-axis gain of sum patterns,

but the + sign will be included throughout to keep the same form to all

the error expressions.

45
z-position of the near-field scanner. For example, a position error

of ± .025A - 10° (@ 3.125 x .008A) leads to a maximum r^ B of about

2 G,rn

± . 1 dB (@ (3.125) x .01) for the center of the main beam.

To date, little experimental data are available with which to

compare the results of eqs . (61), (64) or (67). However, a comparison can be made between eq . (67) and the computations performed by

Rodgrigue, Joy, and Burns [16]. They introduced errors into a hypothetical near-field distribution in order to compute the effects

of the errors on the far-field. The results of their computations

for the effect of random phase errors (or equivalently z-position

errors) on on-axis gain are plotted in figure 3-14 or A-21 of [16] , and are reproduced here by the dashed line in figure 11 below. The

solid line is n^ plotted from eq . (67). One can see from figure G,rn r n v J &

11 that the two curves are in close agreement. The solid line lies

slightly above the computed dashed line as indeed it should if eq

(67) represents an upper bound.

Figure 12, which will be explained in greater detail below

under instrumentation errors, shows a comparison between the effects

on the on-axis gain of a sum pattern from systematic (quadratic in

this case) phase errors introduced into actual near-field data by

Newell [17,21] and the corresponding upper-bound results calculated

from eq. (64). Again agreement is close, with the upper-bound

curve lying just above the actual curve for small deviations in

phase

Next, let's calculate the effect of systematic z-displacement

errors on the depth of the null for an antenna operaing in a difference pattern. From eq . (64)

tv

B

< ± 35 AF 6 s g(r) , 'G,s — max & *• J '

where g ( r ) is no longer equal to unity because r is in the direction

of the boresight null. Typically, AF is about .01 and the depth of

the null is 25 dB down from the main beams of the difference pattern. Thus

g(7) ~- 10 5/4

- 18,

and

n„l < ±b . 3 o G,s — max

46
rn for the null depth. For example, if Az = .01 A. (which would cormax

respond to an accuracy of about .27 dB for the main beams) eq. (68)

insures that the null depth would be accurate to within .40 dB

.

This surprisingly high insensitivity of the null depth to z- displacement errors (or, equivalently , phase errors) has been observed by Newell [21] upon introducing phase errors into the measured

near-field data of a number of antennas operating in the difference

mode. To understand the reasons for this high accuracy in null

depth, one must refer to the derivation in Appendix B. There

appears to be no simple way to explain this result heuristically

.

It is also found in Appendix B that the maximum shift 6 ,

. r . ri^ shift

in the direction of the far-field null of a difference pattern

caused by z-position errors is given by the simple expression which

is not a function of wavelength:

4Az'

max 2X

shift

I

max

tt£ max max

radians

.

(68)

In general, the shift in the null caused by all other sources of

error are negligible compared to this shift caused by the z-position

error or, equivalently, the phase errors.

Table 2 lists the upper-bound position errors in a number of

far-field parameters for a typical X-band and K-band antenna. The

values in the table were calculated from expressions (61) and (68) . As eqs . (61) and (64) show, the maximum possible transverse (xy)

position errors do not depend upon frequency for a given g(r).

Table 2 also shows quite dramatically that the z-position or phase

errors everywhere except in the boresight direction can be extremely

large compared to all other representative sources of error -- compare Tables 1-4. (This is also true of the shift in the difference

pattern null.) Especially note that the error in a -25 dB sidelobe

can be several dB for phase errors (2ttAz /A) of just a few degrees

max

Whether or not these maximum possible far-field errors are actually

experienced in practice depend strongly upon the shape of the nearfield z-position or phase error throughout the scan area. For

example, we shall find in the next section that receiver phase distortion usually has a functional dependence which introduces

47
negligible errors into the sidelobe fields. It is important to

know exactly what effect various distributions of near-field phase

errors have on the far-field in order to avoid experimentally, if

possible, the distributions which produce large far-field errors

in the directions of interest. Such a detailed study of the dependence of the far-field errors on the functional form of the nearfield z-position or phase errors will not be included as part of

this report but will be contained in a forthcoming report by

Newell [21]

2. Instrumentation Errors

The amplitude and phase of the probe output are measured at

discrete points as the probe moves back and forth across the nearfield scan area. The receivers are capable of sampling and recording

the amplitude and phase to within a certain accuracy only. The

errors in the near-field data, caused by the inaccuracies in the

receivers or instrumentation used to measure the probe output, produce errors in the computed far-field. This section estimates the

far-field errors under given limits of accuracy of the instrumentation which measures the amplitude and phase of the probe output. It

is emphasized that the errors produced by the imperfect positioning

of the scanner were determined in the previous section and are not

considered as part of the instrumentation error analysis of this

section. Also the instrumentation errors associated with converting analogue to digital information is assumed negligible.

Under the conditions explained in Section III.B, the far-field

errors can be found in the region

e < x/(icumax

) by evaluating the integral (39)

I = / AE.(P,d) dP.

° A

z o Here, AE (P,d) represents the difference between the measured (E

t

) and "actual" (E ) output of the probe at the point (P,d), i.e.

AE (P,d) = ETM

eas (P,d) - E

t

(P,d). (69)

48
If we look at just the x -component first, the integral I and eq.

(69) combine to give

I ox

= / (E^eaS

-Ex

) dP. (70)

o By writing

and

E x

= |E |-e

X

(71a)

i(cf> +ACJ) 1

) E

meas = M E

i + AA ± ]e x x (71b)

X

L

' X ' X

J ' K J

where AA and Acf> are the errors in amplitude and phase respectively

-A. -A. introduced by the measuring instrumentation, eq. (70) becomes

I« v

= / [Cl'E l+AA^e X -|E |]e x dP. (72) ox i

L v ' x ' x' ' X ' J *• J o

I I In general, both AA and Ac)) are functions of the transverse coordi-

.A. -A. -p

' nates ¥. For small errors A<J) << 1, so that

yC

x

« 1 + iAcf) 1

- Y x— x 2 and

I ox

= / [AA* + i(|E

x

| + AA^)A^ - -^(A^Me X dP. (73)

o All terms higher than second order have been neglected in eq . (73).

icj> x If a sum pattern is assumed and the exponential e is written as

in eq. (47)

,

i<J> x

i(kd+<|> ) e = e (cos A* + i sinA(j) ),

-A. A

with the arbitrary phase constant <J> chosen as in eq. (47), then to

OX

a high probability, the integration in eq. (73) which is multiplied

by the oscillating quantity sinA<j> can be neglected compared to the

-A.

49
maximum possible value of the cosA<j) integration. (Note the disI x tmction between Act and Acj) . A* is the actual variation in phase

of the x-component of electric field within A , whereas Acb is the

o

T x error in phase introduced by the phase measuring instruments.) In

addition, since ( | E | +AA ) cosAcb is always greater than zero for sum A. A. X

patterns, the reference phase for the instrumentation phase error

Acf) can be chosen to make

! (|E

x |+AA^)A^ cosA<t> x

= 0, (74)

o which again expresses essentially the same condition as that chosen

by Ruze [15] in his work on antenna tolerance theory. Under the

above conditions, the imaginary term within the brackets of eq. (73)

is eliminated and the magnitude to of I reduces to

ox

T l E I T |I 0X I = 1/ [AA

x

- ^^(A4>

x

) 2 ]cosAcJ) x dP|

(75)

1{ Kl d¥+ iKmax^ 2 I

l E

xl d¥- o

' o

Acf> ov is the maximum value (in absolute value) of the instrumentation xmax -. n phase error within the partial scan area A . Since it can be shown that (see eq . (10), (30b) or (52))

/ |E I dP = ArlE I (76)

o where |E I is the magnitude of the maximum x-component of farxo r^"°° field, I I I may be rewritten ' ' ox ' ;

Because the errors in measuring phase are usually greatest at points

in the scan area where the amplitude is least, it is desirable to

choose the partial scan area A as small as possible when estimating

Acj) 1

. The far-field within <2A/£max

T is hardly effected by the

xmax ' ' J near-field outside that part of the scan area where the amplitude of

electric field is equal to the edge taper down from the maximum

amplitude. Thus for the sake of designating maximum errors in phase

measurement on a near-field scan plane, the area A need be chosen

no larger than this effective "edge taper" scan area.

50
I I < J I AA1

t dP + tCAcf)

1

) 2 Ar|E | ox ' — i

' x

1 2^ Y xmax ; ' xo

o

p->oo (77)

It is usually possible to express the errors in amplitude in dB

per dB change of amplitude. That is, the receivers are assumed to

read the correct (zero error) amplitude of the probe output at its

maximum value point on the area A . At all other points the amplitude lies a certain number of dB down from its maximum value.

Typically the dB errors in measuring the amplitude are linearly

related to this number of dB down from the maximum amplitude, and

thus the errors can be expressed in dB per dB down. Even if the

actual amplitude error curve is not linear, for the sake of the

upper bound expression (77), it can be replaced by a straight line

(linear curve) which is equal to or greater than the actual error

curve .Specifically, if N, R designates the amplitude error in the

number of dB per dB down, and A JT) the amplitude in dB down from the

j

Old maximum amplitude, AA can be expressed as (for small errors)

AA I"x x

NdB W8 - 7 (78)

By definition

A

dB 20 log xo xo

IE X

(79)

where E denotes the maximum of E on the scan area

xo ' x

' tion of eq. (79) into eq. (78) and the result into eq.

Substitu-

(77) yields

ox

< N

IdB / (E -IE

A

xo

) dP + =-(Acj) ) 2 Ar|E I J 2

V Y xmax ;

' xo ' r-^°° (80a)

With the help of eqs . (57) and (76), eq . (80a) becomes

'aA

I < ox ' — NdB A

- 1 + ^(Acf)

1

) 2

^ T xmax; Ar E

xo ' r-*-°° (80b)

By combining eq. (80b) and the corresponding equation for |l | with

o eq. (50) , assuming E _.-£___, and approximating —r- by 1, the upper-bound

expression for the fractional far-field error n caused by the

51
instrumentation is obtained 11

n(r) < [Nj

B (a-l) ^(A<

ax )

2

] 9 <^max

(sum patterns) . (81)

Equation (81), which is analagous to the position error equation

max (60a), holds only for sum patterns within an angle A/(10£ ) of the

boresight direction. For difference patterns as well, and for

> A/(10£ ), the following upper bound expression analagous to

the position error equation (61) applies:

n(r) < [2N^

B

(3

:

) + n*]^^

(A(J> )

where

dB

8AF A(J)

A<(>

max

r

= <

max

(a-D

2/g(r)

(a-l)/2

(sum patterns)

(difference patterns)

(sum and

difference patterns)

(sum patterns)

e (difference patterns)

A

10£max

10£

A

max

<e<

10£ max

A (sum and difference

patterns)

aALmax

101

10A

max

<e<

2A

max

3A

<e< 1

V. I max

(82)

= wavelength.

= the maximum instrumentation errors involved in measuring

the amplitude of the probe output -- N,„ is expressed in

error per dB amplitude down from the maximum amplitude on

the scan area. (For the present purposes, the amplitude

error is designated as zero at the maximum amplitude.)

Hwhen the maximum value of the probe output on the scan area for the

x-orientation is very different from that of the y-orientation

(E / E ) , it can be shown that eq . (81) remains valid as an upper

yo xo n v bound for the errors in magnitude of the far-field. But for errors

T in polarization ratio an extra term, N , R

| x | /8 . 7, must be added to

eq. (81), where x

p

is the difference between the maximum probe outputs measured in dB for the two orientations.

52
Ad)Y = the maximum instrumentation errors (expressed in radians)

max r J involved in measuring the phase of the probe output on the

effective scan area A (see footnotes 10 and 13)

.

o

J - fractional difference between the amplitude of the two main

far-field lobes of the difference pattern (see Appendix B)

.

'taper" factor -- equal to a minimum of 1.0 (for apertures

AF

a = a

g(r)

max

max

of uniform amplitude and phase) and less than 5 for most

tapered distributions found in practice. (See eq . (31) for

the precise definition of a; for a difference pattern one

should still use the taper factor of the constituent sum

patterns .

)

= ratio of the amplitude of the maximum far-electric-field to

the far- electric-field at the given direction r, i.e., the

inverse of the normalized far-field pattern. (g(r) = 1 for

the center of the main beam, or beams if a difference pattern.)

= maximum width of the antenna aperture.

= maximum width of scan area.

= area of antenna aperture.

The derivation of r] is identical to that done for z-position errors

z

I X 2 X in Section III.B.l. The derivation of 3 for <0< is

io£max

£

max

ax accomplished by the same procedure used above for 9 < X/(10£m ).

In the far sidelobe region, 9 > 10A/&max the far-field errors become approximately equal to the corresponding errors in the nearfield amplitude, and after using eqs. (30b), (31), and (79) 3 in

this region can be written as shown in eq. (82) . Between 9 equal to

may rn o "Y" 2X/£ and lOX/il the value of 3 can be estimated by connecting

rnn v rn o "V" a straight line from its value at 2A/£ to its value at 10A/JI

3 near the boresight direction < X/(10£ ) of difference patterns

is derived in Appendix B. (Note that the error factor 3 is generally

much smaller near the boresight direction or null axis of difference

patterns than near the center of the main beam of sum patterns. This

result occurs, as Appendix B shows, because the instrumentation distorts the amplitude on the "positive" and "negative" sides of a difference pattern by approximately the same amount.) Appendix B also

shows that we can write an upper-bound expression for the null shift

of difference patterns caused by the instrumentation errors:

53
2Ad) X Y max , . u . £ . < radians. shift — n max (83)

Equation (83) is identical to eq . (68) with the instrumentation phase

error replacing the z-position error.

Equation (82) represents an upper bound to the far-field errors

produced by the instrumentation which measures the amplitude and

phase of the probe output. It applies to either sum or difference

patterns in the forward hemisphere of all electrically large aperture antennas. As with eq. (61), eq . (82) was derived for scan

planes which are parallel or nearly parallel to the plane perpendicular

to the electrical boresight direction. (If desired, the errors in

the gain function, sidelobe level, polarization ratio, and beamwidth

may be calculated from eqs . (2) -(5) once n and the far-field pattern

are known .

)

A comparison of eq. (82) with eq . (61) shows that the instrumentation phase error Ad) in eq . (82) has taken the place of 6 eq

in

max l A max

(61). This result acts as a check on eqs. (61) and (82) because

6 simply represents a phase error caused by a z-displacement

max -i -j -p error in the position r of the scanner. Also, Ad) can be separated

max r into a random and systematic part to get a result analagous to eq

(64). However, the random phase errors introduced by the instrumentation are usually much smaller than the systematic phase errors,

and thus can usually be neglected.

Figure 11 reveals that the phase (Acf> ) part of eq. (82) for

max

the on-axis gain of a sum pattern is in good agreement (as an upper

bound) with the computer error analysis performed by Rodrigue, Joy

and Burns [16] on a hypothetical near-field distribution with random

phase errors. Newell [17,21] has introduced phase errors which are

quadratic with respect to the xy coordinates, into the actual nearfield data of the 60 GHz, 46 cm (18 inch) reflector antenna whose

12 Note that the "phase error terms" in eqs. (61) and (82) for sum

patterns and < A/(10£max ) depend on the square of 8 and A6

JIlci.A. IIld-A. respectively. Thus, these two squared terms should not be added

directly when estimating the total error in the far-field. Instead,

6 should be added to Acf) 1 before the square is taken.

max max

54
near-field amplitude envelope is shown in figure 7b. The resulting

JD

errors in r\ r for on-axis gain computed by Newell are shown with the

dashed line in figure 12. The solid line represents the maximum

phase error plotted from eqs . (82) and (2c). Again it appears that

the expression (82) represents a useful upper bound estimate for the

instrumentation phase errors in on-axis gain, especially when the

measured phase is accurate to within a few degrees across the effective

scan area A (as is usually the case in practice).

The accuracy with which the receiver can measure the phase of

the probe output is related to the amplitude of the probe output.

Specifically, the smaller the amplitude the larger the phase errors

usually become (see footnote 10). For example, a typical receiver

used at the NBS near-field range measures phase to within ± .001

radians (± .05°) at the maximum amplitude on the near-field scan

plane, and ± .01 radians (.5°) at an amplitude 20 dB down from the

maximum. Thus, for an edge taper 20 dB down, AcJ) =.01, and eq . (82) max

shows that the error in the main beam of a sum pattern caused by the

errors in measuring the phase of the probe output are negligible

(r^ = Z^L (-01) 2 < ± .001 dB) . The same is true for the null

depth and shift of difference patterns. Of course, the errors in

sidelobe level could be affected to a greater extent by the phase

errors, depending on the shape and distribution of the phase errors

across the scan area. However, since receiver phase errors usually

increase monotonically with decreasing amplitude, it can be shown

as a consequence [21] that the upper-bound off-axis or sidelobe

phase errors (A<j) . /2 g(r)) given in eq. (82) represents a much

larger error in far-field than would usually occur in practice.

Thus , in general , receiver phase errors have a relatively small

effect over the entire far- field of sum or difference patterns . In

fact, compared to the maximum possible effect that typical z-position

errors can have on the off-axis far- fields, instrumentation phase

errors can be ignored completely in the off-axis region.

Even for aperture antennas with edge tapers greater than 20 dB down,

it is unlikely that the near-fields outside these -20 dB points (or

even the -15 dB points) have a significant effect on the maximum possible far- field errors. In other words, regardless of how large the

edge taper, the effective scan area A need not extend beyond about

the -15 or -20 dB points.

55
For receivers which measure phase with high accuracy, the phase

part of the instrumentation error can be neglected and only the

amplitude error remains in eq. (82), i.e.

n(r) < N^Cg 1

) g(F).

Amplitude errors for < 2\/ I do not depend directly on frequency

or the size of the antenna, only on the taper factor a of the nearfield beam, the receiver inaccuracy N, R

, and the inverse of the

normalized far-field pattern g(r). (For null depth of a difference

pattern the amplitude errors are extremely small and do not even involve g(r). These results are proven in Appendix B.) Also, in the

far sidelobe region, 6 > 10A/& , the instrumentation amplitude

errors are relatively small, usually less than a few tenths of a dB

for N, R less than a few thousands of a dB per dB . The far-field

error in the on-axis (g(r) = 1) gain for sum patterns is found from

the above equation and (2c) to be

n£B

< ± 8.7 N*

B (a-l). (84)

Note that for a=l (uniform amplitude distribution) the far-field

error, caused by the instrumentation errors in measuring near-field

amplitude, equals zero--as it should since the receivers measure

essentially at a constant amplitude across the effective scan area.

The maximum error in on-axis gain (eq. (84)), which is linear

with respect to N, R

, is plotted with the solid lines in figure 13 for

different values of a. Rodrigue et al . [16] have also computed

linear amplitude errors for their hypothetical near-field distribution, which has an a exactly equal to 3.0. Their results (see

figures 3-5 or A-5 of [16]) are reproduced by the dotted line in

figure 13. The errors computed by Rodrigue et al . and the maximum

errors predicted by the analytically derived expression (84) are in

good agreement. The solid line for a=3 lies above the dotted line -- as it must if eq. (84) represents a valid upper bound expression for

the errors

To insure an on-axis gain error less than ± .01 dB , the error

in measuring the near-field amplitude should be kept less than about

± .001 dB per dB down. Unfortunately, the accuracy of most receiver

56
systems is of the order of ± .01 dB per dB down (rather than ± .001)

which, according to figure 13, can lead to errors in the on-axis

gain of about ± .1 dB . This value of far-field error can be larger

than the errors from all other sources combined. Thus, if high

accuracy is desired, special effort should be devoted to designing a receiver system which can measure the amplitude of the probe output

to better than ± .001 dB per dB down. Alternatively, the amplitude

calibration curve for the receivers could be determined to within

± .001 dB per dB down and the errors in amplitude compensated for

by including the calibration curve as part of the computer program

that deconvolutes the near-field data.

This latter correction procedure has been adopted by A.C. Newell

et al. at the National Bureau of Standards. The difference between

the on-axis gain computed by Newell [17,21] with and without the ampli

tude calibration curve is shown by the dashed line in figure 13 for

the 46 cm (18 in) reflector antenna operating at 60 GHz. (Actually,

Newell found a .112 dB on-axis gain difference for an amplitude calibration curve that deviated by at most .02 dB per dB down over the

effective scan area. The dashed line in figure 13 assumes linearity

and simply connects the origin to the point .112 dB at .02 dB per dB .

)

The value of a for this antenna was estimated at 2 from the measured

near- field data shown in figure 7b. Again it is seen from figure 13

that the computed errors for this particular antenna correspond

quite well with the maximum possible errors predicted by the general

expression (84) for a = 2.

In brief, the computations of both Rodrigue et al . and Newell

indicate that eq. (82) yields reasonable values for the maximum farfield errors expected from instrumentation errors in measuring the

amplitude and phase of the probe output.

Table 3 lists some representative far-field amplitude errors

calculated from eq. (82) for an antenna with taper factor a equal to

3, and a receiver nonlinearity in measuring amplitude (N, R

) equal to

.002 dB per dB . Note the extremely small effect that amplitude

errors have on the null depth of difference patterns. As Appendix B

shows, this small effect on the null depth is due to the fact that

the receiver which measures the amplitude of the probe output distorts

the opposite sides of the near-field difference pattern by approximately the same amount.

57
C. Multiple Reflections

Consider a probe which scans on a plane in the near-field of a radiating test antenna. The radiation that the probe receives can

be described by an infinite series of rapidly decreasing terms, with

the first term equal to the unperturbed field of the test antenna.

This unperturbed field scatters from the probe, reflects from the

test antenna and other nearby objects, and returns to the probe to

give the second term in the series. The return radiation again

scatters from the probe, reflects, and returns to the probe to yield

the third term. The process repeats ad infinitum.

In order to determine the far-field of the test antenna by

"deconvoluting" the measured near-field data (without knowing the

detailed scattering properties of the probe or test antenna) , the

multiple reflections must be neglected. That is, the second and

higher order terms in the infinite series just described are assumed

negligible when applying the planar near-field scanning techniques [3]

Multiple reflections can be reduced by decreasing the size of

the probe antenna, by increasing the distance between the probe and

test antenna, and by appropriately covering the scanning range with

efficient absorber material. These measures will not, however,

eliminate the multiple reflections entirely. It is the purpose of

this section to estimate the effects of the multiple reflections on

the far-field which is computed from the near-field scan data under

the assumption of zero multiple reflections

As we shall see shortly, it is a fairly simple matter to estimate maximum and minimum values expected for the far-field errors

produced by the multiple reflections. However, it is impossible to

derive an accurate estimate of the far-field errors analytically

without knowing the phase of the multiply reflected fields throughout

the scan area. Thus, the only reliable way to get an accurate estimate of the far-field errors caused by multiple reflections is through

measurement. Specifically, a number of near-field scans could be

14in principle, the effect of multiple reflections could be eliminated

at microwave frequencies by the use of gated sinewaves instead of CW.

Unfortunately, the speed of electromagnetic propagation (c) is so great

that the necessary gating times are too short for the present-day electronics to handle. It is possible, however, in the analagous measurement of electroacoustic transducers to eliminate the problem of multiple

reflections by the use of gated sinewaves because the speed of sound

is much smaller than that of light [18].

58
taken on parallel planes separated by about 1/4 wavelength and the

far-field computed from each scan. Any differences observed in the

far-fields computed from the separate near-field scans would indicate

the extent of the effect of multiple reflections (assuming the scan

area is large enough so that changes in finite scan errors are negligible) . In this way the far-field errors can be determined

straightforwardly and accurately. Of course, the main disadvantage

of this "straightforward method" of determining errors lies in the

time and effort it takes to record data and compute far-fields from

several near-field scan planes for every antenna that has to be measured. Because of this disadvantage it may prove worthwhile, particularly when the multiple reflections are very small, to derive

the following very approximate, yet general, upper and lower bound

expressions for the far-field errors caused by the multiple reflections. In addition to the upper and lower limits of errors, the farfield errors will be derived for multiply reflected fields which

satisfy a certain class of hypothetical near-field distributions.

Consider a probe antenna scanning on a plane in the near-field

of an electrically large, aperture antenna. Assume once again, for

the sake of simplifying the mathematics, that the probe behaves as

an electric dipole, i.e. its output in one orientation is proportional to E and in a second orientation proportional to E . Then

x_ y

the error (AE) in the computed far-electric- field can be expressed

with the aid of eqs . (37) and (12d)

,

AE < "A* e ik(r-dcos6)

j Agnr (F

^e

" 1 rR " P

dF< (85a)

;f->-co

rn "p The superscript "mr" on AE denotes that part of the transverse

near-electric-field caused by multiple reflections, and A' refers

to the scan area.

The determination of (AE) requires the evaluation of the

integral

-i-R-P

I = / AETMr (P,d)e r dP, (85b)

A'

Z

59
or if the x- component is concentrated on first

I = / AE

I ' 1I (P,d)e r dP. (85c) x

The amplitude and phase of AEmr

can be written explicitly as

• ^nir

,pinr . .mr r x AE = AA e x x

to recast eq. (85c) in the form

i(cJ)

mr -^R.P)

I = / AATMre x r dP. (86) x A

, x

The maximum value of the integral in eq . (86) occurs when <p mT

= —R»P,

i.e.

I < / AATMr dP. (87) x A

, x

If polarization is not changed drastically upon reflection from the

mr probe and test antenna, AA will be roughly proportional to the

magnitude of the x-component of electric field at the probe. That is,

Mmr a £

mr |E , } X X ' X ' ' v * mr where e is an average proportionality constant between the amplitude of the multiply reflected x-component of electric field and the

amplitude of the total x-component of electric field as the probe

traverses the scan area. Substitution of eqs . (88) and (52) into

eq. (87) yields

I < e mr

A r IE I . (89a)

x — x ' xo ' r->°°

v

In the same manner, the expression for I is found to be

I < e mr

A r IE I . (89b)

y — y

l yo ' r->°° An upper bound expression for the fractional error n emerges when

t eqs. (89) and (85) are combined with eq . (1):

n(F) < e mr g(F), (90)

60
mr where e is the average proportionality constant between the amplitude of the multiply reflected electric field (probe output) and the

amplitude of the total electric field (probe output) as the probe

traverses the scan area. The value of e can be estimated experimentally by changing the distance between the probe and test antenna

by a few wavelengths at various locations within the scan area.

Periodic variations in the amplitude of the probe output which repeated about every A/2 would be caused primarily by the multiple reflections. As usual, g(r) is the ratio of the amplitude of the maximum far-electric-field to the far-electric -field at the given

direction r. Equation (90) represents an upper bound expression for the errors

in the far-field caused by multiple reflections. It applies to the

entire far-field of electrically large aperture antennas. However,

for nearly all antenna-probe interactions it will give much too large

an estimate of the far-field errors because it was derived under the

unrealistic assumption that the multiply reflected fields possessed

just the right phase across the scan area to maximize the far-field

errors. In reality, as the probe traverses the scan area it will

usually experience large variations in the phase of the multiply

reflected fields throughout the scan area that will greatly reduce

their effect on the far field. We emphasize the word "usually"

because there exist some antennas (like the constrained lens, fixedbeam array show.n in figure 9) which present a rather flat reflective

surface to the radiation scattered from the probe, and thus a rather

constant effective path length and phase for the multiply reflected

fields as the probe scans on a plane parallel to the aperture. In

that case the maximum error could be experienced at the center of

the main beam.

The multiple reflections would have a minimum effect on the

far-field when their phase varied radically over the scan area, or

rn y V

more precisely, when cj) - — R«P in eq. (86) has no critical points

and varies rapidly with position P on the scan area. Then eq. (86)

written in polar coordinates (p, cj) ) can be integrated by parts

with respect to p to give

X x

s 271/

X

3f/dp'

P e P

P' d V (91)

2tt AATMr(p',cj) ) ikf(p\ct )

o

61
where p' refers to the distance from the origin within the scan area

to the point on the boundary of the scan area at the angle $ . The

function f(p',<J> ), which is assumed to possess no stationary points

p

with respect to p, is defined by

kf = <f> mr

x

- y

R ' p - (92)

As mentioned above, the smallest values of I occur, in general,

when the phase function f varies rapidly with p and cf> . Experience

at the NBS [17] indicates that it is unlikely that the average phase

variations in multiple reflections which occur in practice ever

exceed 360° per wavelength across the scan area. Thus it is also

unlikely that | I | will be smaller than its value when f is chosen

as a function which changes an average of about 360° per wavelength

of motion in any direction across the scan area. For example, one

such function is

f = p(l + cos

2 9 ) . (93)

Substitution of the function (93) into eq . (91) shows that the magnitude of I may be expressed as

x

7 ^

>-£

2, AA^p',* ) /

- x— p' e ikp' (l+cos 2 9^)

d(J) l+cos^e

p

(94)

Equation (94) is written as an inequality to emphasize that for

nearly all antennas |l | would be larger than the right side of eq

(94).

In order to get an idea of the value of eq . (94), assume that

<< p' so that the points of stationary phase of p ' (l + cos 2 ^ )

d(J) p

with respect to <b occur near

P

*2l

= 0, tt/2, tt and 3tt/2. Then inte

gration by the method of stationary phase shows that eq. (94) may be

written approximately as

- lis > * /A L

*ve AA

ave (95a)

with L equal to the average of the width of the scan area, and

AAave equal to the average amplitude of the x-component of the multiply reflected electric field at the four points <\> = 0, tt/2, tt and 3tt/2

62
on the boundary o£ the scan area. If we approximate AA by

e |E I XX , where |E | refers to the average amplitude of the xX

component of electric field on the boundary of the scan area, eq.

(95a) becomes

x

A /, , ave mrip i ave —

> t~ vX L e El 2tt x ' x

'

(95b)

Since II = /|I : + | I

| 2

, eq. (95b) combines with the corresponding

x y

equation for | I | to give

y

I > -^ /A L 1 — 2ttave

A yC (96)

The average amplitude |E | of the transverse electric field may be

expressed r in terms of the maximum transverse electric field (E^ ) on

to

the scan plane,

-X

ave

E

t

| ave

= 10 20

'to (97)

.ave with X denoting the number of dB down from that maximum (see

footnote 7)

.

Equations (30b) and (31) can be utilized in conjunction with

eqs . (97) and (96) to give the final expression for the minimum value

of Hi ,

X

ave

I I > /A L

ave

10 20 mr

e a A

2 r I E I (98)

From eqs. (98), (85) and (1) we find the minimum value expected for

the fractional error n in the far-field caused by multiple reflections

X

ave

n(r) >

f A 1 ave

3/2 10 20 mr ,—

.

a £ SO) (99)

(The aperture area A has been taken as . 5(L ) 2 .) The inequality

(99) represents a lower bound expression for n in the sense that the

actual far-field errors caused by multiple reflections would, to a high probability, be greater than but could lie reasonably close to

63
the value of the right side of eq. (99). Of course, there remains

the possibility that for some points in the far-field the effects of

the multiple reflections will cancel to such a degree that n would

actually be less than the right side of eq . (99). These exceptional

points must be ignored if eq . (99) is accepted as a valid lower

bound

Between eqs . (90) and (99) we have approximate upper and lower

limits to the value of n,

X

ave

A 1 ^L

ave

3/2 a 10 20

mr ,—. . ,—, . mr

e g(r) n(r) e g(r) (100)

where

mr

X

a

ave

ave

= wavelength.

= the average ratio of the amplitude of the multiply reflected

probe output to the amplitude of the total probe output

as the probe traverses the scan area. (Its value can be

estimated experimentally by changing the distance between the

probe and test antenna at various locations within the scan

area, and calculating one-half the fractional peak to peak

height of the variations in amplitude that repeat about

every X/

.

)

= average width of the scan area.

= the average amplitude of the probe output at the boundary of

the scan area measured in dB down from the maximum amplitude

of probe output in the scan plane.

= a "taper" factor -- equal to a minimum of 1.0 (for apertures

of uniform amplitude and phase) and less than 5 for most

tapered distributions found in practice (see eq. (31) for the

precise definition of a; for a difference pattern one should

still use the taper factor of the constituent sum patterns)

g(r) = ratio of the amplitude of the maximum far-electric-field to

the far-electric-field at the given direction r, i.e., the

inverse of the normalized far-field pattern. (g(r) = 1 for

the center of the main beam, or beams if a difference pattern.)

64
(If desired, upper and lower limits for the errors in the gain function, sidelobe level, polarization ratio, and beamwidth may be found

by combining eq. (100) with eqs . (2) -(5) and the far-field pattern.)

The value of the factor

„ave

x 1 3/2 a 10

20

T avej TT

for a typical microwave antenna and scan area is on the order of

.001; in which case eq. (100) becomes

.001 e mr g(r) < n(r) < e mr g(r). (101)

It is clear from eq. (101) that the errors in the far-field

produced by multiply reflected fields of a given relative amplitude,

i.e. a given e , can span an extremely wide range of values depending on the variation in phase of the multiply reflected fields across

the scan area. (For example, if the multiple reflections are down

40 dB (e = .01), eqs. (101) and (2c) show that the multiple reflection error in the on-axis gain of the main beam lies between

about ± .0001 and ± .1 dB . ) Essentially, the right side of eq. (101)

gives the far-field errors when the effective phase of the multiply

reflected fields is uniform, and the left side when the phase varies

an average of 360° every wavelength across the scan area. Because

of the extremely large range in the possible value of n, it appears

unlikely that a precise value of the far-field errors produced by

the multiple reflections can be obtained by any method other than

direct measurement. As was mentioned at the beginning of this section, data could be taken on a number of parallel scan planes separated by about A./4, and the far-fields computed for each plane. Any

differences noted in the computed far-fields for the separate scan

planes would be caused primarily by the multiple reflections. In

addition, it appears likely that the effect of multiple reflections

could be reduced appreciably by averaging the far-fields obtained

from a number of different scan planes separated by a small fraction

of a wavelength over a distance of one wavelength.

Finally, we shall evaluate the far-field errors for multiply

reflected fields which are described by a class of hypothetical

65
near-field distributions. In particular, assume that the fields are

mr linearly polarized and that cb in eq . (86) has the functional

dependence

,

cb = 2tt 1

X

2

J (102)

where A, and A~ are arbitrary real constants. Also, assume that the

test antenna has a circular aperture of radius "a" and uniform amplimr tude distribution. Then for scans in the very near-field, AA in

mr

x eq. (86) can be approximated by a constant (AE ) inside the aperture

area and zero outside. Under these conditions, the integral (85b)

may be written in polar coordinates as

,mr

ztt a

1 I^

3 [3—cos 8 +-r—sincb -sin0cos(cb -cb)]

A

L A

I = AE"

11 / / e J J o o

p A pdpdcb (103)

The <b integration can be performed after writing the bracketed

expression in the exponential of eq. (103) in the form

1— cos* + t~ sincb - sin0cos((j) -<j>) = B cos(cb -b)

A, p A

? p p p

(104)

where

and

B = /(A/A^sinecoscb) 2 + (A/ \^- sinBsincb) (105a)

b = tan

t— - sinG sincb

A -, sine cosi

(105b)

Using the integral representation of the Bessel function,

2tt J (z) = J^ / e iz cosip

2tt

dijj

eq. (103) reduces to

,mr

I = 2ttAE

]111 / J

O

J O A o

2TrpB

p dp (106)

66
and eq. (106) becomes

I = AaAEmr 1

2-rraB A (107)

By combining eqs . (107), (85), and (1), the fractional error n in

the far-field takes the form

aAE mr

n(r) = J

1 27raB

A

Br |E(r)

(108)

;£->-oo For a circular aperture with uniform amplitude distribution E , e< (10) can be integrated to yield the far-field,

E(r) I = E - (cos 2 9 + sin 2 cos 2

(j)) J

2Tra

A sin6

cose

(109)

The definition of n(r) loses significance when its value becomes

larger than 1, i.e. when the far-field amplitude |E(r)| becomes

less than the amplitude of the far-field error. To avoid this

9-TT Q

situation, J-, (—r— sine) in eq . (109) can be approximated by its

envelope

,

ffiz -

|J]_(z) I ~ J

x (z) = '

7TZ

\ z

2

) z < 2

z > 2

(110)

Substitution of eq. (110) into eq . (109) and the result into eq. (108)

transforms the expression for n into

mr . „ e sine

n(r) =

(2^aB^

1 X 1

B(cos 2 e + sin 2 e cos

2

4)) J

2-rra sin6

(111)

r mr A -c mr /X2 ^ (e* = AE /E ) o ' o

J

If A, and X

? (see eq . (102)) are chosen such that A, = A? = A , and

(J) is set equal to zero, B can be written from eq . (105a) as

A

^ o sine + sin

67
and r\ becomes

mr - a e smQ

n(6) =

j

2Tra 4-

o t—sineo +sin^0

X

X X

X -sinf +sin 2 6 J^ 2fTa

X

(112)

sin9

We can compare this result with the previous maximum and

"minimum" values estimated for r\ in eq . (100) when g(r) = a = 1, and

X =0. For X = °° (uniform phase for the multiply reflected

fields) eq. (112) should equal the right side of eq . (100). For

X = X eq. (112) should be of the same order of magnitude as the

left side of eq . (100). Indeed, when X eq. (112) becomes e mr

(the right side of eq. (100)), and when A = X it reduces to approxi'

mately .06 (A/a) 3/2

(100) since A = na

2 was taken as .5(L ) in eq

which is nearly equal to the left side of eq.

(100). This

agreement between eqs . (112) and (110) supports the validity of both

expressions

Equation (112) reveals that the on-axis far-field error n(o)

can be expressed in the especially simple form,

n(o) = emr

J

X (113)

Figure 14 shows the on-axis error n(o) plotted against the variable

/2i\a/X . Note that for X < 2a, n(o) is given approximately by

.06 3/2

cos 8.9

X

mr

mr

and never gets larger than about .1 z"

x

. That is, if the phase of

the multiply reflected fields changes 360° or more across the diameter

of the aperture area, the error in the on-axis far-electric- field is

mr mr

less than one-tenth e , where e is the ratio of the amplitude of

the multiply reflected electric field (probe output) to the total

electric field (probe output) as the probe traverses the effective

scan area. Only when the phase of the multiply reflected fields is

uniform (X = °°) across the scan area does the on-axis far-field

o

, mr

error equal e In figure 15, n(6) is plotted for different values of a/X

Q

from

eq. (112) for a/A = 12. When X is greater than a couple of aperture

AC
diameters, the envelope of the errors from multiple reflections are

mr

on the order of e throughout & the far-field. As X becomes less

o than a couple of diameters, the near-axis errors grow much smaller

mr than e but the envelope of errors in the far sidelobes remains at

mr about e . Finally, when X gets as small as the free-space wavemr length A, the far-field errors are much smaller than e all the way

to 6 = 30° in the far-field.

IV. Summary

The far-field characteristics of a radiating test antenna can

be determined throughout the forward hemisphere by scanning on a near-field plane of the test antenna with a probe antenna of arbitrary but known receiving characteristics. The amplitude and phase

of the probe output are recorded on the near-field scan plane and

the far-field pattern is computed by "deconvolving" the near-field

data. The accuracy of the computed far-field depends upon the size

of the scan area, the accuracy with which the scanner positions the

probe, the accuracy with which the instrumentation measures the

amplitude and phase of the probe output, the extent of the multiple

reflections, the computation errors involved with deconvoluting the

near-field data, and, of course, the accuracy with which the probe

is calibrated and the input power to the test antenna is measured.

Essentially, this report derives upper bound expressions for the

errors in the far-field pattern produced by these sources of error

in the near-field measurements. The upper-bound expressions are

written in a form that can be used to stipulate design criteria for

the construction of near-field scanning facilities. In particular,

the limits of accuracy in a given far-field parameter are expressed

in terms of the measured near-field data and/or the computed farfield, the frequency and dimensions of the antenna-probe system, the

systematic and random variation in the positioning of the scanner,

and the precision of the instrumentation which measures the probe

output. In order to simplify the mathematics, the probe was usually

assumed to be an electric dipole, although the resulting upper bound

expressions hold for arbitrary probes.

69
The analysis and resulting upper-bound expressions are not restricted to a particular antenna as previous computer studies [11,16]

and direct far-field comparisons have been, but apply generally to

electrically large aperture antennas which can be operating in either

a sum or difference pattern. The results for position and instrumentation errors apply only to nonscanning antennas in the sense that

the analysis for these two sources of error assumed that the scan

plane lay perpendicular or nearly perpendicular to the boresight

direction of the antenna pattern. Position and instrumentation

errors for beams which are steered away from the perpendicular to

the scan plane will be included in a subsequent report by Newell [21]

Except for the position and instrumentation errors, however, the results of the present report, and in particular the error expressions

pertaining to the truncation of the scan plane, apply to arbitrarily

steered antennas.

Broadbeam antennas where the wavelength is on the same order

of magnitude as the dimensions of the aperture are not examined in

this report. But it is shown in a report by Crawford et al . [20]

that many of the conclusions and upper bound expressions derived

here apply directly or in slightly modified form to broadbeam

antennas

It is emphasized that the upper-bound expressions derived in

this report determine the limits of accuracy of the far-field computed from the planar near-field scanning technique without resorting

to comparisons with direct far-field measurements-. This is probably

the foremost purpose of the report along with the report acting as

an aid in deciding design criteria and tolerances for the construction of new near-field scanning facilities. It has been the feeling

of those involved with near-field measurement techniques at the NBS

that often the near-field techniques determine the far-field more

accurately than conventional far-field measurements with a standard

antenna. Thus comparison with measurements made on conventional

'far- field" ranges would not be a reliable method, even if it were

feasible, for estimating the accuracy of the near-field techniques,

which do not have the problems of proximity corrections, ground reflections, or the need of a standard far-field antenna. A brief

summary of the major conclusions and results of the report follows:

70
Errors in computation can be ignored immediately. A simple

exercise in Section I showed that their effect on the far-field is

extremely small compared to the effects of the other sources of

error. Far-field errors from the approximation involved in applying

the sampling theorem are also negligible (see footnote 1)

.

In Section II it was demonstrated that errors in various farfield parameters could be expressed conveniently in terms of the

fractional far-electric-field error (nvi = I

I AE II'll / I E I ) and the approxi- p-^-oo ' r tr mate far-field pattern. In particular, the errors in gain function,

sidelobe level, polarization ratio, and beamwidth were expressed in

terms of n (see eqs. (l)-(5)).

In Section III. A the far-field errors associated with neglecting

the near-fields outside the finite scan area were investigated.

First it was shown analytically that no reliable information about

the far-field pattern could be obtained by the planar near-field scan

method outside the "solid angle" formed by the edge of the antenna

aperture and the boundary of the scan area. Well within this solid

angle (0 < ~- , see eq. (33) and following), reasonable upper

l mci x bound expressions for the finite scan errors were found that could

be applied to center-line data (eq. (32)) as well as full-scan data

(eq. (36) or (32)). As part of the finite scan analysis, asymptotic expres

sions for the near-fields in front of a circular antenna of uniform

aperture distribution were derived in Appendix A and plotted in

figures A3 and A4 . A comparison was made between the empirical

analysis of finite scan errors performed by Newell and Crawford [11]

with centerline data and the maximum errors calculated from the upper

bound expression (32). Agreement is quite reasonable, as figures 8a

and 8b indicate. Table 1 shows the finite scan error for various

far-field parameters of a typical x

- band antenna. The finite scan

errors are proportional to wavelength, so changing the wavelength

while holding the other antenna dimensions the same merely changes

the values in Table 1 proportionately. Of course, such an isolated

change is rather unrealistic.

Deviations in the position of the probe from its assumed position in the scan area will produce errors in the near-field data

which show up as errors in the computed far-field pattern. Section

71
III.B.l derives an upper bound expression throughout the forward

hemisphere for the far-field errors produced by both systematic and

random errors in the positioning of the probe (see eqs . (61), (64),

(68) and figures 10, 11, and 12). (Of course, a uniform displacement

of the scanner does not alter the far-field pattern, and a uniform

rotation of the scanner simply rotates the entire far-field pattern

through the same angle.) The contribution from the inaccuracies in

the position of the scanner divides naturally into a transverse or

xy (parallel to the scan area) and longitudinal or z (perpendicular

to the scan area) part.

For sum patterns, the on-axis far-field errors were found proportional to the xy-displacement errors of the scanner but proportional to the square of the z-displacement errors (normalized to

wavelength) . This latter result is analogous to that obtained by

Ruze [15] in his classical work on "antenna tolerance theory" for a small arbitrary phase error or aberration in the surface of an antenna. This result does not imply that the z-displacement errors

generally have a much smaller effect on the main beam than the xy

errors, because the multiplying factor is, in general, much larger

for the z term. It does imply, however, that random and systematic

displacement errors in the longitudinal or z direction weigh equally

in their contribution to errors near the center of the main beam of

sum far-field patterns. Unlike the z errors, random errors in the

transverse or xy-displacement of the scanner have a negligible effect

throughout the far-field compared to systematic transverse errors of

the same order of magnitude, and the xy-position errors do not depend

on wavelength. The xy-position errors also differ from the z-position

errors in that the same xy error expression applies throughout the

far-field hemisphere whereas the z errors are given by one expression

close to the main beam, i.e., near the boresight direction, and

another expression for the errors off -axis. Specifically, the maximum possible off-axis (or sidelobe) z-position errors depend linearly

upon the systematic z-displacement errors of the scanner and are

much larger than the on-axis errors, which are proportional to the

square of the z-displacement errors. In fact, as Tables 2 and 4

indicate, these off-axis errors caused by displacements in the

72
z-position of the scanner can be much larger than the total off-axis

errors from all other sources combined. The present report does not

examine in detail the relationship between the distribution of z- displacement errors and the off -axis or sidelobe errors, but such an

analysis will be included as part of the subsequent report by Newell

[21].

For difference patterns, the effect of the near-field displacement errors on the far-field were given by the same upper-bound

expressions derived for sum patterns, except for the effect of z- displacement errors on the null depth. This error in null depth for

difference patterns was shown in Appendix B to be independent of the

value of the null depth itself and proportional to the z -displacement

errors, but, in general, proportional by a very small proportionality

constant. In fact, for most antennas and reasonably accurate scanning systems, the z-displacement errors do not affect the null depth

by more than a few tenths of a dB . This rather surprising result,

which is rather difficult to explain without going through the mathematics of Appendix B, has been confirmed by an empirical error

analysis performed by Newell [21] on the measured near-field data of

a number of antennas operating in the difference mode. Although the

z-displacement errors do not have a strong influence on the depth of

the null of difference patterns, eq . (68), which was also derived in

Appendix B, shows that they do have a strong influence on the direction of this null. In fact, the effect of all other sources of

errors combined on the null direction of difference patterns is

generally negligible compared to the effect of z-position errors,

although instrumentation phase errors can sometimes shift the null

an appreciable amount as well.

Table 2 shows the xy- and z-position error in various far-field

parameters for a typical X-band and K-band antenna. Again, note the

strong influence that z-position or phase errors can have on the

off-axis far-field parameters (sidelobe level, beamwidth, mainlobes

of difference patterns) and on the null shift of the difference

pattern. Note also that the maximum possible null shift caused by

z-position errors is independent of frequency.

73
The far-field errors caused by the inaccuracies of the receivers

in measuring the phase and amplitude of the probe output are estimated in Section III.B.2 (see eqs. (82), (83) and figures 11-13).

(It should be mentioned that instrumentation errors associated with

converting analog to digital information is assumed negligible.)

As would be expected, the instrumentation errors in measuring phase

contribute to the far-field errors in exactly the same way as longitudinal or z-position errors of the scanner. However, since the

errors which a typical receiver introduces into the phase are small

and increase monotonically with decreasing amplitude of the probe

output, calculations show that their effect is often negligible

throughout the far-field.

Although typical instrumentation errors in measuring phase are

small and often introduce insignificant errors into the far-field,

typical instrumentation errors in measuring amplitude can have a pronounced effect on the far-field (see figure 13), except for the

far sidelobe region and in the null depth of difference patterns.

In general, if high accuracy is desired, the receiving system which

measures the amplitude of the probe output should be calibrated and

the calibration curves included as part of the computer program

which deconvolutes the near-field data to get the far-field. It

is significant, however, that the instrumentation errors in measuring

near-field amplitude have a relatively small affect on the null depth

of difference patterns, as eq . (82) and Table 3 demonstrate. The

reason for this is that the receiver distorts the opposite lobes of

the near-field amplitude by approximately the same amount (see

Appendix B) . Table 3 also displays the amplitude error in various

other far-field parameters for a typical microwave antenna. The instrumentation amplitude errors for 8 < 2X/£ do not depend directly

upon frequency or the size of the test antenna, only upon the taper

factor a of the near-field amplitude, the receiver inaccuracy N, R

, and, except for the null depth of difference patterns, the inverse of

the normalized far-field pattern g(r). It should also be pointed out

that the instrumentation amplitude errors in the far sidelobe region,

> 10X/ i

max

, are relatively small, usually less than a few tenths

of a dB for N, R less than a few thousands of a dB per dB

74
Whenever comparisons were possible, the expressions for both

position and instrumentation errors (eqs. (61), (64), (68), (82),

and (83)) agreed well as an upper-bound with the results of the

empirical error analysis of Newell et al . [11,17,21] at the National

Bureau of Standards, and with the error analysis performed by

Rodrigue et al. [16] at the Georgia Institute of Technology with a hypothetical near-field distribution (see figures 11-13).

In Section III.C upper and "lower" bound expressions were

derived for the far-field errors caused by multiple reflections (see

eq. (100)). The upper and lower bounds were extremely far apart

because the phase of the multiply reflected fields has a strong influence on the far-field errors. Since the phase of the multiply

reflected fields for a given test and probe antenna interaction

would, in general, be difficult to measure or estimate, it was concluded that the only reliable way to get an accurate estimate of the

far-field errors from multiple reflections would be to use the following straightforward but tedious procedure. Take several nearfield scans on parallel planes separated by about 1/4 wavelength or

less. Any deviations in the far-field patterns computed from the

data on the separate scan planes would be caused primarily by the

multiple reflections (assuming the scan area is large enough so that

changes in finite scan errors are negligible). If necessary, the

effect of multiple reflections could probably be reduced appreciably

by averaging the amplitude of the far-fields obtained from these

different scan planes separated by a small fraction of a wavelength

and covering a total change in separation distance of one wavelength

In addition to the upper and lower limits of error, the far-field

errors were derived for multiply reflected fields which satisfy a certain class of hypothetical near-field distributions (see eq.

(112) and figures 14 and 15)

.

The major analytical results of this error analysis study can

be combined into one long upper-bound expression for the fractional

far-electric- field error n(6,(|0. (6 and <J> specify the angular

spherical coordinates of the far-field pattern.)

75
n(e,cj))

, T max, n aAL 10 20 aAA'

max

A sine

n probe

max

11max

2N dB

(3l)

+ 2 mr

2

input

null shift of 2A

shift difference pattern — max

7T£

6

S

+A<j>

X max T max radians

(114a)

(114b)

(6 max a<kLj 2 + max (6

rn

v ) max ;

8AF(6 S +A0 1

v ) max Y max ;

r

B

1 -<

max

(o-l)

2/g(e,<fr)

(a-l)/2

, . max aAL

3A

(sum patterns)

(difference

patterns)

(sum and difference

patterns)

(sum patterns)

(difference

patterns)

(sum and difference

patterns)

10Amax

10Amax

10£

max

A

10£

10A

max

<e<

2A

,max

£

max

<e< I

Equations (114) represent essentially an amalgamation of eqs . (32),

(64), (68), (82), (83) and (100), under the extreme condition that

the various near-field errors combine in such a way as to create the

maximum possible far-field errors. If desired, the tighter upperbound, eq . (36), for the finite scan error could be used for the

first term in eq . (114a) instead of eq. (32). Recall from Section

III. A. 3 that the finite scan error term represents a valid upper-bound

halfway or more within the "solid angle" formed by the edges of the

aperture r and the boundary' of the scan area (9v <

"• _ 9 = ~-(90-y ) ; and

2 max 2

K 'max'

outside this solid angle region the planar near-field scanning technique cannot be relied upon with any confidence to yield accurate

far-fields. The instrumentation amplitude factor 3 is not given

explicitly by eqs. (114) in the region between 9 equal to 2A/£

and 10A/£ . However 3 can be estimated in this region by connecting a straight line from its value at 2X/1 to its value at 10A/£

The detailed derivation of each of the terms in eq. (114), except

76
n , and n. . , can be found in the part of the main text from probe input r which each particular r term came. The extra terms n , and n. probe input

will be explained below. The definition of the various parameters

in eqs . (114) can also be found from the preceeding main text:

X = wavelength.

A = area of the antenna aperture.

TH £L X

I = maximum width of the antenna aperture.

L = maximum width of the scan area.

y

= maximum acute angle between the plane of the scan area

'max & r and any line connecting the edges of the aperture and

scan area (0 = 90-y )

.

^ max 'max'

X = the largest amplitude of the probe output at the edge of

the scan area, measured in dB down from the maximum

amplitude of probe output in the scan plane.

a = a "taper" factor- -equal to a minimum of 1.0 (for apertures

of uniform amplitude and phase) and less than 5 for most

tapered distributions found in practice. (See eq. (31)

for the precise definition of a; for a difference pattern

one should still use the taper factor of the constituent

sum patterns .

)

A = 2ttAP /A, where AP is the maximum amplitude of the

max max ' max r transverse (xy) displacement errors within the effective

scan area Ao^o . (A is that part of the scan area over

^ which the phase is fairly uniform. For near-field scans

parallel to the aperture A - A.)

8 = 2ttAz /A, where Az is the maximum amplitude of the

max max ' max r longitudinal (z) displacement errors within the effective

scan area A . I °

AcbY = the maximum instrumentation errors (expressed in radians) max y r j involved in measuring the phase of the probe output on

the effective scan area A (see footnotes 10 and 13)

.

o AF = fractional difference between the amplitude of the two

main far-field lobes of the difference pattern (see

Appendix B)

.

Njt, = the maximum instrumentation errors involved in measuring

the amplitude of the probe output --N, R is expressed in dB

error per dB amplitude down from the maximum amplitude

77
on the scan area. (The amplitude error is designated

as zero at the maximum amplitude; see footnote 11.)

mr

z = the average ratio of the amplitude of the multiply reflected probe output to the amplitude of the total

probe output as the probe traverses the scan area.

(Its value can be estimated experimentally by changing

the distance between the probe and test antenna at various locations within the scan area, and calculating

one-half the fractional peak to peak height of the variations in amplitude which repeat about every A/2.)

g(9,<j)) = the ratio of the amplitude of the maximum far-electric

field to the far-electric -field at the given direction

(6,(f)), i.e., the inverse of the normalized far-field

pattern. (g(9,<j>) = 1 for the center of the main beam,

or beams if a difference pattern.)

The superscripts "s" and "rn" refer to the "systematic" and "random"

parts of the displacement errors respectively. The instrumentation

phase and amplitude errors (A<J> , N, n ) are assumed systematic in

I max ai5 i max

nature. The phase error Ac}) does not show up in n for 9 > A/(10£ ) in cix. z because, as explained in Section III.B.2, the shape of the near-field

instrumentation phase error is such that it has negligible effect in

this off-axis region compared to the maximum effect of typical syss tematic z-position r errors 6 . Random position errors have nonmax v negligible effect only near the boresight direction of sum patterns.

mr The multiple reflection ratio e is enclosed in a box in eq.

(114a) to emphasize that for most antennas it represents an unrealistically large upper bound.

The error n , (9,cb) simply represents the uncertainty in the probe v ,rJ r j r receiving characteristics of the probe in the direction corresponding

to (9,(J)). For example, if the receiving characteristic S'-. of the
probe (see reference [3]) was known to an accuracy of 1% for the

direction (9,6), k. > r j y then n , would be .01 for that direction. 'probe

The error n . . arises in normalizing the amplitude of the input r probe output to the input power or amplitude |a | of the input mode

to the test antenna. Such a measurement is necessary whenever absolute values of the gain function are required. Probably the simplest

and most accurate method of performing this normalization is to connect the input waveguide of the test antenna directly to the output

waveguide of the probe through a variable attenuator. If, as explained in Section III.B.2, the receiver which measures the amplitude

of the probe output is specified arbitrarily to have zero error when

the probe output is at its maximum amplitude on the scan area, then

the normalization can be accomplished by measuring the attenuation

needed to reduce the amplitude of the direct input from the transmitter to the level of the maximum amplitude of the probe output on

the scan area. Of course, "mismatch factors" of any consequence

must also be measured. The quantity n 7 n. . merely denotes the com- input J bined fractional error of the variable attenuator and of the devices

used to measure the necessary mismatch factors. By using a high

precision r attenuator,' the fractional error n. . can usually be input ' kept below a few thousandths.

Table 4 shows the total maximum possible error in a number of

far- field parameters for a typical X-band and K-band antenna and a reasonably accurate scanning facility. The table was computed from

eqs. (114) for the representative values of the near- and far-field

parameters listed above the table. It is emphasized that the values

shown in Table 4 are the maximum possible upper-bounds to the farfield errors computed under the extreme condition that each of the

sources of near-field error produce its maximum possible change in

the far- field and then all these maximum changes in the far-field

add in phase. The z-position errors have been separated and underlined when they represent the dominant contribution to the upperbound errors, because whether or not these maximum possible z-position

errors actually occur depends strongly on the far-field direction of

interest and on the shape of the deviation in z-position throughout

the scan area. As mentioned above, a detailed analysis of this dependence will be included as part of a subsequent report by Newell

[21].

79
The error Cn be

+ ^

ut ) in the receiving characteristic of

the probe and in normalizing to the input power of the test antenna

was chosen as . 1 dB in Table 4. For the error in on-axis gain of

the sum patterns, Table 4 shows that this contribution of .1 dB is

about as large as all the other errors combined. The same is true

for the gain of the mainlobes of the difference pattern if the z- position part of the errors is ignored. Thus, for the situation

described by Table 4, greater accuracy in the calibration of the

probe and in the measuring of the input power to the test antenna

could be a first step in significantly reducing the errors in the

direction of maximum gain.

It is interesting to compare the maximum equivalent reflected

signal [24] allowable in a conventional "far-field" range or anechoic chamber to get accuracies comparable to those shown in Table 4

for the near-field scanning technique. It is a simple matter to

show that the maximum equivalent reflected signal (ERS) measured in

dB down from the direct signal is related to the values of ri, R by

n dB ERS = 20 log -X—~ (assuming small ri,J .

For example, near the boresight direction the ERS would have to be

33 dB down for the sum pattern and 15 dB down for the difference

pattern to give the corresponding errors shown in Table 4. Of course,

there would also be errors on conventional ranges due to proximity

effects, uncertainties in the calibration of the standard antenna,

and instrumentation errors.

In conclusion, it can be seen from Tables 1-4 that for a reasonably accurate scanning system no one source of error dominates over

all the others in the boresight direction of both sum and difference

patterns. However, in the far-field region away from the boresight

direction, but well within the solid angle region formed by the edge

of the aperture and boundary of the scan area, the deviation in z- position of the scanner can, in principle, cause far-field errors

which are much larger than the combined errors of all other sources.

In practice, however, these maximum possible errors seldom occur.

Moreover, it will be shown in the subsequent report by Newell [21]

80
that the scanner can be designed and utilized to keep the off-axis,

z-position errors far below the upper-bound values given by eqs

.

(114) and shown in Tables 2 and 4. Of course, beyond the solid angle

region formed by the edge of the aperture and boundary of the scan

area, it was shown in Section III. A that the far-field computed from

the near-field data cannot be relied upon with any confidence.
Ef-2 E AE

Figure 1. Main beam of a hypothetical test antenna,

Aperture of Test Antenna

»- z

Figure 2. Schematic of scanning geometry

82
Test Antenna

Figure 3. Schematic of aperture antenna,

Figure 4. Definition of a , 6 , D . 6 m m m

83
P.Pm Dm /j

m* point

/ rm

r^-*]

t

"Xem

R ^\9

>o'm

»- 0'

j rl 0' %

Figure 5. Schematic of aperture and scan areas. (Although

A and A' are drawn parallel, it is not a necessary

requirement of the theory.)

e=tan~'(-/2-l)

Reflector

Figure 6. Circular reflector antenna.

84
-5

-10

GO-aS

" l5

<u -oB -20

"5.

<f -25

<-o2-30

o

a>>-35

UJ -40

-45

-80 -60 -40 -20 20 40 60 80

Probe X- position in cm

Figure 7a. Near-field centerline data (constrained lens)

(z = 25 cm)

.

-50 -40 -30 -20 -10 10 20

Probe X-position in cm

50

Figure 7b. Near-field centerline data (reflector antenna)

(z = 43.18 cm).
.15

CD

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Q>o>cD

t_ o

c

oo

</> Xo

c

" .05 o

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Upper

bound

\

\

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Crawford

2.6 24 2.2 2.0 1.4 1.0 (L /2a) -Scan Length /Aperture Diameter

Figure 8a. Change in gain vs. decreasing scan length (constrained lens)

CD

c

o

.co

c

"o

CO

co

.25

20

10

.05

.05

10

15

.20

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i

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i

s

2.05 1.85 1.65 1.45 1.25 1.05 .85 .65 45 .25 .05 (Lmox/2a)- Scan Length /Aperture Diameter

Figure 8b. Change in gain vs. decreasing scan length

(reflector antenna)

86
Figure 9b

Figure 9a. Constrained lens sum port near- field log amplitude, £=9.2 GHz, z-25.0 cm, no radome.

Near-field phase,

constrained lens sum port,

f = 9.2 GHz,

z = 25 cm.

Figure 9c. Constrained lens antenna and probe

87
288

252

<f> 216

d)

CDwm

<v IHO o

c

a; 144

ino

si

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Figure 9d. Near-Field Centerline Data, z = 25 cm

88
±1.4

±1.2

±1.0

±0.8

±0.6

±0.4

±0.2

Figure 10. Position errors in on-axis gain.

±.5i—

±10 ±20

Figure 11. Comparison with Rodrigue et al. for random phase

errors.

89
±.25

±20

CO "3 ±15 o

Ic m

P- ±10

±.05

Upper

bound

Newell

mm ——— —

±1 ±2 ±3 ±4 , ±5 ±6 ±7 ±8

Degrees (±A<£ max)

Figure 12. Comparison with Newell for quadratic phase errors.

±07

±06

±.05

±04

CD

±.03

±.02

±.01

a=3.0

y a=2.5

/Rod

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rigueetaL

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a =2.0 X /

V

Newell

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/

a=l.5

Jr *"

* X

±.001 ±.002 ±.003 ±.004 ±.005 ±.006 ±007 ±008 ±009

N^

B (dB per dB down)

Figure 13. Amplitude errors in on-axis gain.

90
Figure 14. On-axis error in far-field from multiple reflections.

a/X =I.O

I.Oc

env elope

\ -K

—

0° *2>

6°r<«12° h18° 24°r30° 6 a/X =8.0

Figure 15. Far-field errors from multiple reflections

(a/A = 12)

.

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i—i CD+ 1 P

CD

+-» CDg

03P

a3 Oh

T3

i—1 CD •H

P<1 P

o3

Pin

PQp

•H

as

taO

co •HX

aSi PO

PQ

r-1 >

CD

i—l

CD XiO

i—1

CD •Hm o

-e-

+-> • H6

a3

CD PQ o• H

+-> o3PPo• H

4-> o3HP

03 t—1 O

d,

t/1 • HX

03i PO

PQt3M

CD

i—l i—lD2

coCD

CDPM

CD

+-> <-H •H

CO

i—I

i—lP2

PQ-P

t/1 CDO

i—lP

•H

o3 2 PP 6 CD 3 -M

CO 4-» a3Ph

CDup

p

CD P P CD

CD -M m +-> m o3 H P,

03PP

CD PP<p

03 PQp

03 X

03U

•H

PUXH

aSPo

COPoppwpo

COoI P

a3

>-Xo

I—I

03

E-i

93
OQ

Tiu

CDCu

CQT3 T3

rvi CD i_n O rj

ro • i

II II II a DQ i—

t—1 H^J CD z >

CD

CD X>O

i—

CD

H3)

CO ouUW

<d

+-> H

i—

e<

cq

LT) too

+i

COX)

to+ 1 c

-e- o

CD+1 OO+ 1

CQ

-CO- LO

CD+ 1

CD r—

X>

•H

DOH

t—

DC

CD Z

X)

COH

CD

CD6

03

Jh

CC

13

i—

CD •H

PL,

Jh

03

LL,

H

03

en •HX<1 o

.—1CD>

CD

CD X)O

r—

CD T3

•H

CO o

-e4-> -0

•H>6

03

CD CO o

•HM

o3 QiCO

.—i M

03

tsj H

J-i 03 i—

O

D_

'/) •HX<1 co M

CD Q

r—

I—

z M

4-1 •H

CO

r—

r—z

CDqc e E CD

C/) +-> 03

CDU

u

J-H CD

CD +-> U-t 4-> ch o3 •H CX

T3 o<

DCcHa

03

CD

'7) J-o

in PJ

o3

4-> c

CD£

Jh

4->

[fl C

tO

CD i—

-C

03 H

94
CQ CQ -d T3

1—1

CQT3

LO o I—1 O

<xi ro CD o T—1

co

rH

X PL,

u +J

+-> < e a > CO ea d J Q t3 -h

CU r—I +

_o rH CD o 3 x>

r—1 2 o

CD CQ 5_, 13) TJ P,

•H CD

p:e

CQ

CDp

CQT3

CQ

X E

X) 03 CO E E o

LO CU rH o CD

<=t o • r—1 •

II II II II II X X X X CQ

03

d o3 03 I—i-C>

-/. e f-i E h£ z M rj e- < <3 < E u O EU o o o o ^o o o CD

to 1—1oo LO VD

II II II II II a X X < X

03 03 03 E E

CD o

-e- E ,—s U , N t-~ LO r-« / V to • CD /—>

ro vD «3- • vO to • + + CD • rH CD • +

oo rH rj •=t + *-h

II CD • O to r~- i—1

i—i

!h CXI • CD . CD • O r< • v—

v—

• rH v_* v—

Jh

f* +1 +1 + 1 + 1 + 1 + 1 + i PUrH

03 o

4-> -e- O E ,—» H u ,—,

oo

LO to , N vO • o ' v t~-

(XI (XI . vO I—1 • + + CD • rO VXD • +

O0 1—1

en CD + *3"

II 00

rH • CD CXI CD

LO CM r-i

<-< • '—"

v

—

• -—1 * p ^ + 1 +1 +1 + 1 + 1 +1 + 1 o

•H

4-> 03

i-> 5-, CU

+->

d /•

v CU / N O (O E CQ •H CU

03 ,-

—

N

T3) +-> CU

fH PQ

v

—

03 S-. 03 X N , s tie Dh *—

rH •H PQ CU ,—s

CU rH T3 "d PQ t3 fi > o 03 v—

v—

Xi

rH •H CU e- r—1 v

—

CU 03 I—1 o .d +-> •H bJD x P +-> <-H (0 PU CU M P. •H CU I to x> T3 to CU H X)

5-. • H o •H •H T3 CO o

03

X rH £ X I—1

PL, 03 CU E 03 rH r—i c i xs 03 i rH r—1 • H c •H CU d 3 3 rt o en PQ o Z z "'•

CUu

d

d c

!-H ca Jh E cu f-H CU P +-> CU +-> CO +-> Mh -m

03 Mh oj a, •H a\

03cc

cu Md<d

03 CQd

03 X

o3U• H

p-X

E03HO

M-t

(O

!hOH

LU

T3

PL,i

03 •MO

CU rH

03

95
Appendix A

Near-Fields of a Circular Antenna with Uniform Aperture Distribution

The purpose of this Appendix is to derive expressions for the

near-field of a circular antenna of uniform aperture distribution.

Specifically, the amplitude and phase are constant within the aperture and zero outside. The antenna is electrically large in the

sense that the diameter of the aperture is many wavelengths across.

The transverse electric field to the right of the aperture is

given exactly by eq. (16). If we assume that the aperture fields are

linearly polarized in the x-direction, eq . (16) becomes

E (r) = x

v J 2tt 9 z

-E . 2tt a iklr-R'

O _8 r r Q

l_ a 7 i J . _ _ , R'dR'dtJ)' (Al)

o o | r-R '

for the circular aperture of radius a and uniform amplitude E . The

element of area has been written in terms of the polar coordinates

(R',cf)') of the vector R'. In addition it can be proven from eq. (15)

that if the aperture fields have longitudinal components which are

small compared to the transverse components, then the longitudinal

components remain relatively small in the near-field (z << a

2 /A) , provided a/A >> 1.

The change of variable t = R'-R, where R is the transverse part

of r, converts eq. (Al) to

E (r) x

v J -2iE

o 3z

ikz 7T ik/t 2 (a) + z

2

/ e da

o R<a (A2a)

E (r) = x

v J 2iE

o dz

ai

o

ik/t 2 (a)+z 2 ik/t 2 (a) + z

2 da R>a (A2b)

The variables t , t, and t and the limit of integration a, are

o 1 2 l

defined in figures Ala and Alb. Apparently, eqs . (A2) were first

derived by Schoch [19]

For large k, the integrals in eqs. (A2) can be evaluated by the

method of stationary phase. The integrand of eq. (A2a) has two

stationary points, one at a = and one at a = tt. The two terms in

96
the integrand of eq. (A2b) each have a single stationary point at

a = 0. After carrying through the details of the method of stationary phase, eqs . (A2) become

E (r) = E [e

lkz +AI] R<a

x

v J o

L J

E x

(r) = E

o AI R>a,

where AI is given by

AI = ^ M * 2tt R

i [k/z 2 +(R+a) 2 -£] i [k/z 2 + (R-a) 2 +J]

(R+a)y^ 2 +(R+a) 2 (R-a) /z 2 + (R-a) 2

(A3a)

(A3b)

(A3c)

Equation (A3b) is identical to eq . (17) of the main text when the

following substitutions are made:

D

1 = /z 2 +(R+a) D

2 = /z 2 +(R-a) 2

a.

cosO-, = z/(R+a) cos6~ = z/(R-a)

Strictly speaking, eqs. (A3) are valid only as k = 2tt/A approaches

infinity. For finite k they represent first term approximations to an

infinite asymptotic series, and the first term approximations are

valid only in certain regions of the near-field. These regions of

validity can be estimated by returning to eqs. (A2) . First concentrate on eq. (A2a) . In eq. (A2a) t ranges from (a-R) to (a+R)

.

Thus if the rate of change& of t is somewhat uniform with a on the

o interval a = to it, the method of stationary phase will yield a good

approximation when

/(R+ a) 2 + z /(a-R) 2 +z 2 >> 1. (A4)

Unfortunately t does not change uniformly with a from a = to it,

but it does so in the separate intervals to y and y to it. Since

t

2 = a2 -R 2 at a = y, the condition (A4) must be replaced by the two

conditions

,

97
/(R+a) 2 + z

2 A2 R2 + z

2 > 3 (A5a)

, , , , /a 2 -R 2 +z 2 - /(a-R) 2 +z 2 > 3. (A5b)

(Actually the right side of eqs . (A5) should be >> 1 but experience

with the method of stationary phase indicates it remains a good

approximation down to where the quantities in brackets in eqs. (A5)

are just a few wavelengths -- specif ically 3A is chosen in eqs. (A5).)

Manipulation of eqs. (A5) shows that they are satisfied for

large a/A if

3A < R < [a- (12A/a)A] (A6a)

z < ^y /R 2 -(3A) 2. (A6b)

That is, eq. (A3a) approximates eq . (A2a) in the region defined by

eqs. (A6) . A similar analysis with eq . (A2b) shows that it is

approximated by eq. (A3b) in the region defined by

R > [a+(12A/a) A] (A7a)

a

z < (R-a) 3A' (A7b)

The regions of validity defined by eqs. (A6) and (A7) are shown

in figure A2 for an a/A equal to 12. Each region is labeled by the

equation which approximates the field in that region. In addition

to the regions defined by eqs. (A6) and (A7) two other regions--one

near the z-axis,

R < 7a

R < the larger of 2a

z or 3(^)

2/3

A

(A8a)

(A8b)

and one near the edge of the aperture

are shown in figure A2

z > 2 r a

z >

Tra a-R

R-a

1

< 4 a

(A9a)

(A9b)

(A9c)

98
In the axis region defined by eqs . (A8) , the exponent in the

integrand of eq. (A2a) can be approximated by the first two terms

of its expansion in R/a, and integrated to give

E (r) = E x

v J o ikz ze

i k /z 2 + a

: J

/z 2

ka

R

^2+a 2 ^

(A10)

z"+a'

Similarly, in the edge region defined by eqs. (A9) , the eq. (A2b)

can be integrated approximately to give

E (r) = J x

v J 2 ikz ze ik/z 2 + 2a 2 J

/z 2 + 2;

ka : Vz 2 + 2a 2 ^

(All)

In the near-field in front of the aperture (z less than about

Yy a

2 /A) , eqs. (A3a) and (A10) reveal that the electric field can be

approximated by a single equation,

with

E (r) = E [e

lkz

+ AI ' ] , x

v J o

L J ' (A12a)

f i[k/z 2 +(R+a) 2 -J] ( AI za

/2

y

{ (R+a)/z 2 +(R+a) 2

i[k/z 2 +(R-a) 2 +|]

ka

R

•/z^+CR+a) 2

(R-a)/z 2 +(R-a) 2

ka

R

Vz 2 + (R-a)

and J defined as

o

J (z) = o

v J

(1 - i z

2

) z < 1.46

z > 1.46.

(A12b)

(A12c)

Essentially, eqs. (A12) represent the total electric field for the

linearly polarized circular aperture in the region

R < a-A (A13a)

1 a

2 z < 12 A*

(A13b)

99
The amplitude |E | and phase 6 of E v may be found from eqs. CA121

,

1 —[Q! cos(k/z 2 + (R+a) 2 -kz-|) - Q 2 cos (k/z 2 + (R-a) 2 -kz + }1

a E

(A14a)

kz za

/2

coskz Q 1 sin(k/z 2 + (R+a) 2 -^) - Q sin(k/z 2 + (R-a) 2 - J)

sinkz Q 1

cos(k/z 2 +(R+a) 2 -|) - Q

2 cos (k/z 2 + (R-a) 2 +j)

with

(A14b)

J' ka

/z 2 +(R+a) 2

R

(A14c)

(R+a)/z 2 + (R+a) 2

Q? =

T e ka

«j _ pK

u /z 2 +(R-a) 2 (A14d)

(R-a)/z 2 +(R-a) 2

Equations (A14) hold (to a first order approximation) everywhere

in the near-field region defined by eqs. (A13) . Except within a wavelength or so of the z-axis, eqs. (A14) reveal that in this region

1 X the maximum amplitude fluctuations are about ± — / — , and the phase

varies slightly from kz. The amplitude of the electric field given

by eq. (A14a) is plotted in figure A3 for a/A = 12 and 15, and for

increments of z/a from to y~ a/A. The amplitude curves show large

fluctuations within a wavelength of the points of maxima and minima

along the z-axis. These z-axis maxima and minima, which are a wellknown phenomenon for the circular aperture of uniform distribution,

are less pronounced for noncircular apertures, or for apertures with

a tapered distribution (see reference [13], Section 1-F).

Figure A3 also shows that the amplitude variations across the

aperture repeat about every wavelength in the very near-field

(z < a/4) , but spread out as the distance from the aperture gets

larger. This behavior has been observed for tapered distributions

as well, by the many near- field measurements on microwave antennas

performed at the National Bureau of Standards [17]

100
The phase in the near-field of a circular antenna with a/A = 12

is plotted in figure A4 . Except near the zeros in on-axis amplitude,

the phase across the beam is uniform to within a few degrees of

oscillation which repeat about every wavelength in the very near-field.

101
(a) b)

Figure Al . Definition of t and a.

Aperture

Figure A2. Dotted line shows region in which eq . (A12a) holds

102
A/\ f\s\P, r

r (z/a = .75)

\j

.2 .4 .6 R/a

w\A/S. S"\. /\ /

\J V

) -- .

vv

(z/a = .5)

' \/

c) .; > .4 .( R/a

3 .8

(z/a = .25)

.2 .4 .6 R/a

|EX

I

p rw\ A A /\ z'

n

^V

(z/a = 1.25)

2 4 6 R/a

(z/a* 1.0)

r^sA?\P\/

2 .4 6 R/a

IE,

A/\ f\~riA>k / 1/v^

(z/a= .75)

vy

.2 .4 .6 R/a

IE J

\i'>^*V/I A fl A /I A

n

U/Q-- 5)

^/v

.2 .4 .6 R/a

(z/a = .25)

.2 4 6 R/a

(a) a/A = 12 ^

a/A = 1S Figure A3. Near-field amplitude of circular antenna. 103
77

*.

.2 .4 .6 .8 R/a

(z/a=.5)

77

*.

-7T

2 .4 R/a

(z/a = .25)

6 .8

IT

+.

77" .2 .4 6 .8 R/a

(z/a=I.O)

77

*.

-77 .2 .4 .6 .8 R/a

(z/a=75)

Figure A4 . Near-field phase of circular antenna (a/A = 12).

104
Appendix B

Position and Instrumentation Errors for the

Null Depth of Difference Patterns

1 . Position Errors

The errors in null depth of a difference pattern caused by

transverse displacement errors (AP) of the near-field scanner can be

derived in the same way and are given by the same upper bound expression as that of the transverse errors for sum patterns (eq.

(60a)). However, the sum pattern derivation of Section III.B.l for

longitudinal displacement errors cannot be applied to finding errors

in the null depth of difference patterns. The main reasons for this

are twofold. The first and most obvious is that the equation for

difference patterns analogous to eq . (48) may not be satisfiable because the near- field of a difference pattern changes phase by 180°

across the aperture. Secondly, as the analysis below shows, the

greatest effect of a longitudinal displacement error is a slight

shift in the position of the null rather than a change in the depth

of the null.

If the longitudinal position errors were zero and the perpendicular (e ) to the scan plane were parallel to the null axis, the

far-field in the direction of the "null" is given in terms of the

near-field by eq. (10):

null m1 e ik(r-d)

j

p- d dp

- (B1)

A

o where, as usual, A refers essentially to that part of the scan area

that covers the antenna aperture, and just the x-component of

electric-field will be considered first.

Now if the amplitude of E (P,d) on opposite halves of the aperx ture (and thus A ) were equal and the phase difference exactly 180

the field in the direction of the null axis would actually be zero.

In reality the amplitudes are not exactly equal and the phase difference, even on the average, is not exactly 180°. Specifically,

we can write E (P,d) as

2v

105
Ex

(P,d)

i(<j> +kd)

(A +AA1 e (over one half, A ,)

A. A 01

A e x

i(<f> ^ r +kd+Aip ) ox r x' (over second half, A

2

)

(B2)

with AAXX/A and Axp both << 1. For simplicity <f> has been chosen X ox

constant, although it can be shown that the results do not change

significantly if cf> is allowed to vary slightly across A . Substi- 0X

iAip ° tution of eq. (B2) into (Bl) and approximating e x by (1 + iAip ) yields

,null

i(kr+c|) ) r y Y ox^ i

Xr A

AA dP

x ol A

/ iA

x

Aij, x dP

o2

(B3)

for the field in the null direction. Experimentally, it has been

found at the NBS that for many if not all antennas operating in a difference mode the first integral in (B3) predominates [17] , but

for the sake of the error analysis we must retain both integrals.

When longitudinal displacement errors (Az) are introduced, eq

(10) shows that eq . (Bl) must be replaced by

.null 1 x Ar

i(kr+(J) ) ik(Az-sin9eD »P)

ox

/ E fP,d) e K dP,

A

X

o

(B4)

where E represents the null field computed from the actual nearfield data containing errors in the z-position of the scanner. 6 can

no longer be set equal to zero because the error Az may shift the

angular position of the null axis. After substituting eq. (B2) into

(B4) , expanding the exponential in a power series, and discarding

error terms higher than second order, eq . (B4) becomes,

106
.null'

J x

i(kr+cf> ) ox r

Ar

/ AA dP - / iA Aip dP

A^i A , X X

ol o2

(B5)

+ / (±A +AA

+

) A

X X

o

ik(Az-6eD 'P) + k

2 Az0ep *P

(kAz) 2

_ Ck9eR -P)* ,

2 2

'R

dP

R

k / A

x A^x (Az-9e

R

-P) dP

A

o2

with

AA in A , x ol

AA

in A

o2

and the + and - sign before A being used in the areas A , and A ~ 6 x

to ol o2

respectively.

The first order error term in eq . (B5) can be made zero by

choosingto 0=6 such that

s

/ (±A

x+AAx

) Az dP =

s

/ (±A

x +AAx )(e

R

-P) dP

o o

(B6)

corresponds essentially to the shift in the direction of the null

caused by longitudinal position errors. If we take the weighted

i — i III £LX average value of [£*P| at about I /4, eq. (B6) implies that for

most antennas

I max

< Az

max '

,max where I denotes the maximum width of the antenna aperture

defining 6 = 2ttAz /X, eq. (B7) becomes approximately,

max max

(B7)

By

(B8)

With the choice of given by eq. (B6) the amplitude,

,null ' r,null i x E"" | , of the error field in the null direction may be

written by subtracting eq. (B3) from eq. (B5)

,

107
pnull ' pnull

x x <h k

2 / ± A

A

x Az0 e n *P

s R

(Az) Ce

s e R

.P) 2 dP

k / A

x

AiP x (Az-6

s e R

.P) dP

o2

(B9)

The first term in the first integral of eq . (B9) can be made zero by

merely shifting the reference plane from which Az is measured. The

third term in the first integral is identically zero because it is

an odd function over the scan area A

o

c null ' ^null i . k x x ' — Ar

Thus eq . (B9) reduces to

k

/ A A^j, (Az-6 |P|) dP + * / ± A (Az)*

A , X X S Z A

X

o2 o (BIO)

The second integral in eq. (BIO) is also negligible if we assume the

rms value of (Az) is approximately equal on the "positive" and "negative" sides of A . Finally, since the maximum value of is 26 A/tt£, J s max

ma)

eq. (BIO) becomes

c null' c null

x x

ave

k 6 A max

Ar I max / A Aij, |P| dP

A

X X

o2

x a i a ve

o Alp

max x

Ar

/ A dP,

A , X

o2

(Bll)

with AiJj denoting the average phase difference from it radians

of the probe output between the positive and negative sides of the

partial scan area which is perpendicular to the null axis . The factor t— / A dP is approximately equal to the maximum

Ko2

field in the mainbeams of the difference pattern. And since an ex- pression analagous to eq. (Bll) holds for the y-component of the

field, we can write the fractional error ri(r) near the null of a difference pattern as

^

1

6

max

A ^ave

^r

)

' (B12a)

where, as in the main text, g(r) is the ratio of the amplitude of the

maximum far-electric-field to the far-electric-field at the given

direction r. Here r is essentially the direction of the null axis

and thus g(r) the ratio of maximum far-field to null axis far-field.

Now eq. (B12) is a very simple expression. However, the average

phase difference A^ of the probe output may be a difficult

108
quantity to estimate accurately from the near-field phase data. Fortunately, it can be shown that Aip is related to the relative size

of the two mainlobes in the far-field of the difference pattern.

Specifically, a straightforward but rather lengthy manipulation

(which will not be shown here) of the near-field of a difference pattern reveals that Ail»r can be approximated by

ave ^

r J

Ail;y - 4AF

ave

where AF is the fractional difference between the amplitude of the

two main far-field lobes of the difference pattern. For example, if

the amplitude of one mainlobe is 10 on some linear scale and the

amplitude of the other mainlobe is 10.1 then AF would equal (10.1-10)/10

or .01. Equation (B12a) can now be written in the alternative form

^

i 46

max

AF

^'

(B12b)

2 . Instrumentation Errors

The inaccuracies in measuring near-field phase have the same

effect on the null-depth as the longitudinal position errors. That

is, A<J> , the maximum instrumentation error in measuring phase,

simply replaces 6 in eqs . (B8) and (B12) . IT13.X The inaccuracies in measuring amplitude affect the null depth

differently, however, than the transverse position errors because

the nonlinearities in the instrumentation distort the amplitude on

each side of the difference pattern in nearly the same way and thus

their effect on the null depth is much smaller than might first be

expected. Specifically, errors in measuring amplitude show up in the

null depth through distortion of only the AA part (see eq. (B2)) of

the near field. Carrying through an analysis similar to that performed above for z-position errors yields the following expression

for the maximum change AE. in null electric field caused by amplitude errors:

|AEnull| 1

_d|

; |AA|) dp £

dB max ol

? (B13)

A

ol Xr

109
where AA is the maximum of AA across A , and N JT. is the maximum

max ol dB

instrumentation error in measuring the amplitude of the probe output

(see Section III.B.2). If we approximate |E. |, the x-component

AAmaxA j of which is given in eq. (B3) , by ^ , then the fractional

z Ar

error in null depth caused by instrumentation errors in measuring

amplitude can be written

(B14)

Note that the instrumentation amplitude error in null depth does not

depend on the null depth itself. In addition, it can be shown that,

unlike phase errors, the instrumentation amplitude errors have

negligible effect on the angular position of the null direction.

110
References

[1] IEEE Standard Definitions of Terms for Antennas , IEEE Transactions on Antennas and Propagation, AP- 22 , 1 (January 1974).

[2] DeSize, L.K. and J.F. Ramsay, in Microwave Scanning Antennas , Ed. R.C. Hansen, Academic Press--New York, London (1964), Vol. 1,

Ch. 2-II.B.

[3] Kerns, D.M., "Correction of Near-Field Antenna Measurements Made

with an Arbitrary But Known Measuring Antenna," Electronics

Letters, 6, 11, pp. 346-347 (28th May 1970).

[4] See e.g. Jackson, J.D., Classical Electrodynamics , John Wiley §

Sons Inc., New York, London (1962), Section 9.6.

[5]a Braunbek, W. , "Zur Beugung an der Kreisscheibe , " Zeitschrift

Fur Physik, 122, pp. 405-415 (1950).

b Ufimtsev, P. la., "Approximate Computation of the Diffraction of

Plane Electromagnetic Waves at Certain Metal Bodies," Soviet

Physics--Technical Physics, 2, 8, pp. 1708-1718 (August 1957).

[6]a Keller, J.B., "Diffraction by an Aperture," Journal of Applied

Physics, 2^, 4, pp. 426-444 (April 1957).

b Keller, J.B., "Geometrical Theory of Diffraction," Journal of

the Optical Society of America, 5_2_, 2, pp. 116-130 (February

1962)

.

[7] Van Kampen, N.G., "An Asymptotic Treatment of Diffraction

Problems," Physica, 14, 9, pp. 575-589 (January 1949).

[8] Keller, J.B., R.M. Lewis, and B.D. Seckler, "Diffraction by an

Aperture II," Journal of Applied Physics, 2_8, 5, pp. 570-579

(May 1957)

.

[9]a Kouyoumjian, R.G., and P.H. Pathak, "A Uniform Geometrical

Theory of Diffraction for an Edge in a Perfectly Conducting

Surface," Proceedings of the IEEE, 62, 11, pp. 1448-1461

(November 1974) . b Hwang, Y.M. and R. G. Kouyoumjian, "A Dyadic Coefficient for an

Electromagnetic Wave Which is Rapidly-Varying at an Edge," URSI

1974 Annual Meeting, Boulder, Colorado.

[10] See e.g., Born M. , and E. Wolf, Principles of Optics , Pergamon

Press, Oxford, New York (1970), 4th Ed., Appendix III.

Ill
[11] Newell, A.C. and M.L. Crawford, "Planar Near-Field Measurements

on High Performance Array Antennas," NBSIR 74-380, National

Bureau of Standards, Boulder, Colorado (July 1974).

[12] Kraus , J.D., Antennas , McGraw-Hill, Inc., New York, Toronto,

London (1950), Chs . 3 and 12.

[13] Hansen, R.C., (Ch. 1, Sect. 2-1 in ref [2]): A /A = G/G in

Table IX, p. 66.

[14] Rusch, W.V.T. and P.D. Potter, Analysis of Reflector Antennas , Academic Press, New York - London (1970), Section 2.52.

[15] Ruze, John, "Antenna Tolerance Theory -- A Review," Proceedings

IEEE, 54, 4, pp. 633-640 (April 1966).

[16] Rodrigue, G.P., E.B. Joy, and C.P. Burns, An Investigation of

the Accuracy of Far-Field Radiation Patterns Determined from

Near-Field Measurements , Report -- Georgia Institute of

Technology, Atlanta, Georgia (August 1973).

[17] Newell, A.C, private communication, National Bureau of Standards, Boulder, Colorado.

[18] Kerns, D.M., "Scattering-Matrix Description and Near-Field Measurements of Electroacoustic Transducers," The Journal of the

Acoustical Society of America, 5_7, 2, pp. 497-507 (February

1975)

[19] Schoch, V.A. , "Betrachtungen liber das Schallfeld einer

Kolbenmembran," Akustische Zeitschrift, 6, pp. 318-326 (1941).

[20] Crawford, M.L., A.C. Newell, J.W. Greene, and A.D. Yaghjian,

"Experimental Design Study for a Near-Field Broadbeam Antenna

Pattern Calibrator," AFAL-TR- 75-179 , Air Force Avionics Lab.,

Wright-Patterson Air Force Base, Ohio 45433 (to be published).

[21] Newell, A.C, "Planar Near-Field Measurement Techniques on High

Performance Arrays -- Part II," Air Force Technical Report, Air

Force Avionics Laboratory -- Wright Patterson Air Force Base,

Ohio (to be published)

[22] Newell, A.C, R.C Baird, and P.F. Wacker, "Accurate Measurement

of Antenna Gain and Polarization at Reduced Distances by an

Extrapolation Technique," IEEE Transactions on Antenna and

Propagation, AP-21 , 4, pp. 418-431 (July 1973).

112
[23] Kanda, M., "Accuracy Considerations in the Measurement of the

Power Gain of a Large Microwave Antenna," IEEE Transactions on

Antennas and Propagation, AP-23 , 3, pp. 407-411 (May 1975).

[24] Appel -Hansen, J., "Reflectivity Level of Radio Anechoic Chambers,"

IEEE Transactions on Antennas and Propagation, AP-21 , pp. 490-

498 (July 1973) . [25] Jensen, Frank, "Electromagnetic near- field-far-field correlations," Ph.D. Dissertation, Technical University of Denmark,

Lyngby, Denmark (July 1970).

ACKNOWLEDGMENT

The author is grateful to Allen C. Newell for his invaluable

consultation throughout the preparation of this report. Much

appreciation is extended to John .W. Greene and Douglas P. Kremer

for the information they provided about the NBS near-field scanning

facilities. Mark T. Ma and Carl F. Stubenrauch of the Department

of Commerce, and Kenneth Grimm of the Air Force Avionics Laboratory

contributed many helpful suggestions in reviewing the report. Many

thanks also go to Janet R. Jasa for carefully and patiently typing

the manuscript

.

113
U.S. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET

1. PUBLK A 1 ION OK HI- PORT NO.

NBS Tech Mote 667

2. Ciov't Accession

No.

4. TITLE AND SUB' Upper-Bound Errors in Far-Field Antenna Parameters

Determined from Planar Near-Field Measurements,

Part I : Analysis

3. Recipient's Accession No.

5. Publication Date October 1975

6. Performing Organization Cod< 276.05

7. AU rHOR(S)

Arthur D Y a g h j i a n

8. Performing Organ. Report No. TN-667

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NATIONAL BUREAU OF STANDARDS,

DEPARTMENT OF COMMERCE

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Bou 1 der Labs

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Air Force Avionics Laboratory

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Air Force Systems Command

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C overed 7/73 - 7/74

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15, SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most significant inforniation. If document includes a significant

bibliography or literature survey, mention it here.)

General expressions are derived for estimating the errors in the

sum or difference far-field pattern of electrically large aperture

antennas which are measured by the planar near-field scanning technique. Upper bounds are determined for the far-field errors produced

by 1) the nonzero fields outside the finite scan area, 2) the inaccuracies in the positioning of the probe, 3) the distortion and nonlinearities of the instrumentation which measures the amplitude and

phase of the probe output, and 4) the multiple reflections. Computational errors, uncertainties in the receiving characteristics of the

probe, and errors involved with measuring the input power to the test

antenna are briefly discussed.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons)

Antennas; error analysis; far^field pattern; near-field measurements;

planar scanning; plane wave spectrum.

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