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NATL INST OF STANDARDS & TECH R.J.C.

A1 1102616271

Cruz, J. E/Assessment of error bounds fo QClOO .U5753 NO.1300 1986 VI 98 C.I NBS-P NBS TECHNICAL NOTE 1300

U.S. DEPARTMENT OF COMMERCE/National Bureau of Standards
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OiOiOO

Assessment of Error Bounds for Some ,,^u

Typical MIL-STD-461/462 Types of tl

Measurements

J.E. Cruz

E.B. Larsen

Electromagnetic Fields Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

Boulder, Colorado 80303

Sponsored by

U.S. Army Aviation Systems Command

DRSAV-QE

St. Louis, Missouri 63120

i>^*' '^o.\

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued October 1986
National Bureau of Standards Technical Note 1300

Natl. Bur. Stand. (U.S.), Tech Note 1300, 108 pages (Oct. 1986)

CODEN:NBTNAE

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1986

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402
Contents

Page

List of Tables v List of Figures vi

1. Overall Introduction 1.1

2. Generating a Known EM Field in a Parallel-Plate Transmission Line... 2.1

2.1 Introduction 2.1

2.2 TEM Cell with a 50 f^ Impedance, 50 Q. Termination,

and 377 fi E/H Ratio 2.2

2.3 Parallel-Plate Line with Open Sides and 377 a E/H Ratio 2.5

2.3.1 Characteristic Impedance and Input Impedance

of a Stripline 2.6

2.3.2 Field Level and Uniformity in a Stripline 2.9

2.3.3 Effect of Screenroom Resonances on Field Strength 2.10

2.4 Operation of a Parallel -Plate Line in the Open-Circuit Mode

or Short-Circuit Mode 2.11

2.5 Use of E- and H-Field Probes as Transfer Standards 2.12

2 . 6 References 2.13

Tables 2-1 to 2-8 2.15

Figures 2-1 to 2-14 2.23

3. EM Field of a Single-Wire Transmission Line in a Screenroom 3.1

3.1 Introduction 3.1

3.2 Experimental Configuration of a Typical Long-Wire Antenna 3.2

3.3 Evaluation of a Long-Wire Antenna 3.3

3.3.1 Field Uniformity Along Axis of Long-Wire 3.3

3.3.2 Measured vs Calculated Field Beneath the Long-Wire 3.4

3.4 Conclusions and Recommendations 3.6

3.5 References 3.6

Figures 3-1 to 3-7 3.7

4. Calibration of Antenna Factors for EMI Antennas 4.1

4.1 Introduction 4.1

4.2 Measurement of Antenna Factors in a Screenroom 4.5

4.3 Measurement of Antenna Factors at an Open Field Site 4.6

4.3.1 Standard Antenna Method 4.6

4.3.2 Standard Field Method 4.8

4.4 Experimental Measurements of Antenna Factors 4.10

4.4.1 Biconical Dipoles 4.10

4.4.2 Vertical Monopoles 4.12

4.5 Discussion of Antenna Factor Measurement Accuracy in

Shielded Enclosures 4.13

4.6 Error Bounds of Screenroom Measurements 4.14

4.7 Conclusions 4.14

4.8 References 4.15

Figures 4-1 to 4-19 4.17

5. Measurement of Broadband Impulsive Signals 5.1

5.1 Introduction 5.1

5.2 Calibration of an EMI Receiver and Measurement of an Impulse Generator Using the Video Pulse Technique of

MIL-STD-462 5.2

5.3 Measurement of an IG by the NBS FFT Calibration System 5.6

5.4 Approximations of IG Output in Terms of the Measured Area

of the Pulses 5.8

5.5 References 5.10

111
CONTENTS (Continued)

Page

Tables 5-1 to 5-4 5.11

Figures 5-1 to 5-6 5.15

6. Overall Sumtnary, Conclusions, and Recommendations 6.1

6.1 Field of a Parallel-Plate Transmission Line 6.1

6.2 Use of an Electrically Small Transfer Probe 6.2

6.3 Field of a Single-Wire Transmission Line 6.2

6.4 Calibration of Antenna Factors for EMI Antennas 6.3

6.5 Measurement of Broadband Impulsive Signals 6.5

7. Acknowledgments 7.1

TV
List of Tables

Page

2-1. Measured input impedance of the NBS MIL-STD-462 stripline

in an unshielded laboratory, with 83 n termination 2.15

2-2. Measured input impedance of the NBS MIL-STD-462 stripline

in an unshielded laboratory, with 120 a termination 2.16

2-3. Measured input impedance of the NBS MIL-STD-462 stripline

in center of NBS screenroom, with 83 Q termination 2.17

2-4. Measured input impedance of the NBS MIL-STD-462 stripline

in center of NBS screenroom, with 120 ^ termination 2.18

2-5. Measured characteristic impedance of MIL-STD-462

stripline in NBS screenroom 2.19

2-6. Measured E-field error in center of MIL-STD-462 stripline 2.20

2-7. Measured E-field error along center line of MIL-STD-462

stripline, at 30 MHz 2.21

2-8. Calculated cavity -mode resonances for the empty NBS

screenroom, first four modes 2.22

5-1 Response of an ideal receiver to various types of

input signals 5.11

5-2 Calibration of receiver impulse bandwidth (IBW) and

impulse generator spectrum amplitude S(f) using

MIL-STD-462 technique, bands 1-4 5.12

5-3 Calibration of receiver impulse bandwidth (IBW) and

impulse generator spectrum amplitude S(f) using

MIL-STD-462 technique, bands 5-8 5.13

5-4 Spectrum amplitude of the commercial impulse generator,

measured with the NBS FFT calibration system 5.14
List of Figures

Page

2-1. Isometric sketch of a typical TEM cell used to generate

standard E and H fields 2.23

2-2. Side view of a typical TEM cell 2.24

2-3. Cross-sectional view of a TEM cell 2.24

2-4. Isometric sketch of a typical MIL-STD-462 parallel-plate

t ransmi ssi on 1 i ne 2.25

2-5. Cross-sectional view of the NBS MIL-STD-462 parallel -pi ate

transmission line 2.25

2-6. Instrumentation for measuring the E-field level in the

MIL-STD-462 stripl i ne 2.26

2-7. Measured input impedance of the MIL-STD-462 stripline in

the center of the NBS screenroom 2.27

2-8. Measured characteristic impedance of the MIL-STD-462

stripline in the NBS screenroom 2.27

2-9. Measured E-field error in center of stripline. Position A

in screenroom, with no ground strap 2.28

2-10. Measured E-field error in center of stripline. Position B in screenroom 2.28

2-11. Measured E-field error along length of stripline

at 30 MHz, 83 Q. termination. Position A in screenroom,

with no ground strap 2.29

2-12. Measured E-field error along length of stripline at

30 MHz, 100 fi termination. Position B in screenroom 2.29

2-13. A parallel -pi ate cage system for generating a known

E-field, 0.01 to 40 MHz 2.30

2-14. Typical response of an EFM-5 probe vs frequency, at a field strength of 20 dBV/m (10 V/m) 2.31

3-1. The long-wire antenna in a shielded enclosure 3.7

3-2. Signal generator drives intermediate feeder line to

long-wire antenna 3.8

3-3. Signal generator coaxial line feeds long-wire directly 3.8

3-4. Electric field uniformity of long-wire antenna along

the length of the chamber at 1 MHz 3.9

VI
Page

3-5. Electric field uniformity of long-wire antenna along

the length of the chamber at 15 MHz 3.9

3-6. Electric field uniformity of long-wire antenna along

the length of the chamber at 30 MHz 3.10

3-7. Calculated-to-measured field versus distance below

the long-wire for selected frequencies from 1 to 30 MHz 3.10

4-1. Test setup required by MIL-STD-461A for determination

of antenna factors 4.17

4-2. Instrumentation used at NBS open-field site to

calibrate monopole antenna factors.. 4.17

4-3. Antenna factor of a typical biconical dipole taken

from the manufacturer's manual for a far-field

separati on di stance 4.18

4-4. Antenna factor of a typical biconical dipole taken

from the manufacturer's manual for a separation

distance of 1 m 4.18

4-5. Antenna factor of a typical biconical dipole taken from the

manufacturer's manual for a separation distance of 3 m 4.19

4-6. Antenna factor of a typical biconical dipole taken from the

manufacturer's manual for a separation distance of 15 m and

an antenna height of 3 m 4.19

4-7. Antenna factor of Biconical No. 1 for a "far-field"

separation distance, as measured at NBS 4.20

4-8. Antenna factor of Biconical No. 2 for a "far-field"

separation distance, as measured at NBS 4.20

4-9. Placement of biconical antennas in the NBS screenroom,

position 1 4.21

4-10. Antenna factors for biconical dipole calibrated in the

NBS screenroom at spacings of 1 and 3 m 4,22

4-11. Placement of biconical antennas in the NBS screenroom,

position 2 4.23

4-12. Placement of biconical antennas in the NBS screenroom,

position 3 4.24

4-13. Measurement results of three screenroom placements with

an antenna separation distance of 1 m 4.25

vn
Page

4-14. Difference between antenna factors measured in a screenroom

and on the NBS field site for a biconical dipole 4.26

4-15. Difference between field strength measured with a calibrated transfer probe (EFM-5) and that measured

with a biconical antenna calibrated by the MIL-STD-461A

two-antenna method 4.26

4-16. Antenna factor for typical 1.04 m vertical monopole

taken from the manufacturer's manual 4.27

4-17. Free-space antenna factor of two vertical monopoles as calibrated at the NBS open-field site 4.27

4-18. Antenna factor for 1.04 m vertical monopole calibrated

in the NBS screenroom at different antenna spacings 4.28

4-19. Antenna factor for 1.04 m vertical monopole calibrated

in the NBS screenroom at different antenna locations for

aim antenna separation 4.28

5-1. Instrumentation for testing the MIL-STD-461/462 approach

for measuring impulse bandwidth of a receiver and spectrum

amplitude of an impulse generator 5.15

5-2. Photographs of the receiver impulse response envelopes

(video pulses) measured at NBS 5.16

5-3. Instrumentation for the NBS automatic FFT pulse

measurement system 5.17

5-4. Photograph of sampling oscilloscope trace of the IG

output pulse (top) and plot of the pulse supplied to

the computer for the FFT (bottom) 5.18

5-5. Graphs of the IG spectrum amplitude measured by two methods,

MIL-STD-462 and the NBS FFT pulse measurement system 5.19

5-6. Sketches of the spectrum amplitude of three common

pulse waveforms 5.20

vm
Assessment of Error Bounds for Some Typical

MIL-STD-461/462 Types of Measurements

J. E. Cruz and E. B. Larsen

Electromagnetic Fields Division

National Bureau of Standards

Boulder, Colorado 80303

This report deals with the instrumentation and equations for

several systems used by the U.S. Army for electromagnetic com- patibility (EMC) testing and calibrations. Most testing for

MIL-STD-461/462 is performed in a shielded enclosure (screenroom)

rather than an open field site, which leads to uncertainty in the

measurement of emissions from electronic equipment, or the susceptibility of equipment to radiation. Assessment of error bounds by

the National Bureau of Standards (NBS) is covered in this report,

and suggestions are given for improving the measurements.

Four areas of concern were studied as follows: (a) electromagnetic (EM) fields generated in a parallel-plate transmission

line (stripline), (b) EM fields beneath a single-wire transmission

line in a screenroom (long-wire line), (c) determination of antenna

factors for electromagnetic interference (EMI) antennas located in

a screenroom, and (d) calibration of EMI receivers to measure

broadband impulsive signals. Most EMC antennas at NBS are calibrated at an open field site or in an anechoic chamber. This

report presents antenna factors determined in a typical screenroom

using the two-antenna method, and comparison with those determined

at an open field site. The video pulse technique prescribed in

MIL-STD-462 for calibrating EMI receivers was also evaluated. Four

different methods were tested for comparison with the MIL-STD

approach. They are defined and discussed in this report.

Keywords: antenna factor; electromagnetic compatibility; field

strength standards; impulse spectrum amplitude; military standards;

screenroom measurements

1. Overall Introduction

This report deals with the instrumentation and equations for several systems used by the U.S Army for electromagnetic compatibility (EMC) testing and

calibrations. The report relates to Military Standards 461, 462, and 463

which are standards issued by the U.S. Department of Defense (DOD) for use by

all departments and agencies within DOD. MIL-STD-461A is titled "Electromagnetic interference characteristics, requirements for equipment." It is

1.1
dated August 1, 1968, and supersedes MIL-STD-461. MIL-STD-462 was issued by

DOD on July 31, 1967, and is titled "Electromagnetic interference characteristics, measurement of." MIL-STD-463 was issued on June 9, 1966, and is titled

"Definitions and systems of units, electromagnetic interference technology."

This report (NBS Tech. Note 1300) deals only with MIL-STD-461A and

MIL-STD-462, and will generally be referred to here as MIL-STD-461/462.

Most EMC testing for MIL-STD-461/462 is performed in a shielded enclosure

(screenroom) rather than at an open field site because of lower cost,

shielding from ambient fields, and convenience. This MIL-STD, which is used

in the procurement of many military and other government systems, has several

recognized problems. Possible errors and large uncertainties can occur in the

measurement of electromagnetic (EM) emissions from electronic equipment, or the susceptibility of this equipment to EM radiation. The uncertainty is

increased when performing the measurements inside a screenroom where multiple

reflections and antenna impedance perturbations occur. Assessment of the error bounds by the National Bureau of Standards (NBS) is covered in this

report, and the data should provide insight into the problems associated with

this type of testing. A few suggestions are also given for improving the

measurement procedures prescribed in MIL-STD-462. However, these improvements

are now being quantified and will be covered in greater detail in a future

report.

Four areas of concern were studied at NBS for determining the error

bounds of MIL-STD-461/462 testing, as follows: (a) EM fields generated in a parallel -pi ate transmission line (stripline); (b) EM fields beneath a singlewire transmission line in a screenroom (long-wire line); (c) determination of

antenna factors for electromagnetic interference (EMI) antennas located in a screenroom, and (d) calibration of EMI receivers to measure broadband

impulsive signals.

The parallel -pi ate stripline parameters (input impedance, characteristic

impedance, and field uniformity) were measured at various locations of the

stripline inside the NBS screenroom, using MIL-STD-462 techniques. The MILSTD also describes a laboratory method of producing a field having known E and

H magnitudes in a shielded room by a long-wire antenna. The objective of NBS

testing was to measure the field level and uniformity, and compare this with

1.2
theoretically calculated values. Equations for computing the E- and H-field

magnitudes in the enclosure are included in this report.

Most EMC antennas at NBS are calibrated at an open field site or in an anechoic chamber, using the standard antenna or standard field method.

Because these antenna factors are not necessarily applicable to making

measurements in a screenroom, the errors for measuring emissions with the

calibrated antenna can be quite large. This report presents antenna factors

determined in a typical screenroom using the two-antenna method, and comparison with those determined at the NBS open field site. Experimental data show

the variability of antenna factor as a function of frequency and location in

the screenroom, thereby providing an indication of error bounds.

The video pulse technique prescribed in MIL-STD-462 for calibrating EMI

receivers was also evaluated at NBS. This approach involves application of a train of identical repetitive pulses to the EMI receiver. As specified in the

MIL-STD, the area and maximum amplitude of the receiver response envelope are measured. From those measurements the receiver impulse bandwidth (IBW) and

spectrum amplitude of the impulse generator (IG) are determined. The IBW of

an EMI receiver is used for measurement of broadband interference. It can be

determined by several different approaches. The spectrum amplitude of an impulse generator as a function of frequency, S(f), can also be measured by

several approaches. Four different methods were evaluated at NBS for comparison with the MIL-STD approach. They are defined and discussed in this report.

1.3

2. Generating a Known EM Field in a Parallel-Plate Transmission Line

2.1 Introduction

Chapters 2 and 3 of this report deal with the instrumentation and design

equations for several systems used to generate calculable electric (E) and

magnetic (H) fields for EMC testing. These "standard" electromagnetic fields

with known magnitude are used to (a) test the susceptibility of electronic

equipment to EM fields, (b) calibrate E- and H- field probes for measuring and

mapping fields, and (c) expose biological specimens in a known EM environment.

The type of instrumentation employed at NBS to generate a standard field

depends on the signal frequency, desired field intensity, and size of the

device under test (DUT). In general, transverse electromagnetic (TEM) cells

or other TEM transmission lines are used at frequencies up to about 200 MHz,

A series of four waveguide transmission cells can be used at frequencies

between 100 and 750 MHz. At frequencies above 200 MHz, NBS usually generates

a known calibrating field in an anechoic chamber, using the transmitted field

of an open-ended waveguide launcher or a pyramidal horn antenna. However, the

production of radiated standard fields in anechoic chambers is described

elsewhere [2-1,2-2] and will not be covered in this report. Most of the EMC

testing for MIL-STD-461/462 is still performed within a shielded enclosure

(screenroom). The screenroom is preferred to an open site because of its

lower cost, shielding from the effects of interfering (ambient) fields, and

its overall convenience--especially during inclement weather.

All of the standard field generators discussed here have been in use for

about 50 years. However, some of the newer EMC engineers dce not familiar

with the principles and operation of these devices. Therefore it seems useful

to summarize the design parameters and mathematical basis for several common

E- and H-field synthesizers.

Much has been written in the past 15 years about the design, construction, and use of so-called TEM cells. Therefore, little discussion is

included here about these TEM lines when operated in their usual mode and terminated in their characteristic impedance, which is usually 50 n. The main

advantage of TEM cells is that electrically-small antennas such as E- and Hfield probes can be calibrated in a shielded environment, in a fairly uniform

EM field having a free-space wave impedance (Z^) of 377 Q, One disadvantage

of TEM cells is the inverse proportionality between their physical size and

2.1
upper useful frequency limit. Another limitation is the large power (and

consequent cost) required to generate a high field intensity, for the usual case where the cell characteristic impedance (Z^) and terminating load

resistance (Z|_) are only 50 ^. Therefore, it is often advantageous to use other field generating configurations for EMC applications.

The following types of EM field simulators are described in chapters 2 and 3 of this report:

(1) TEM cell with a characteristic impedance (Z^) of 50 Q, operated in its

normal mode with a terminating load impedance (Z|_) of 50 q (discussed

only briefly);

(2) parallel-plate transmission line (stripline) with a characteristic impedance (Zj.) of 75-200 fi, but having a free-space wave impedance (Z^) of 377

Ji, where Z^ is the E/H field strength ratio;

(3) stripline or TEM cell operated in the open-circuit mode to produce an intense E-field with a high wave impedance or in the short-circuit mode

to produce an intense H-field with a low wave impedance; and

(4) "long-wire" antenna chamber, made in a typical screenroom by installing a horizontal wire as the "center" conductor of a TEM line, having a I^

value of 200-600 Q,

2.2 TEM Cell with a 50 j^ Impedance, 50 Q, Termination,

and 377 o. E/H Ratio

A TEM cell is essentially a triplate transmission line with the sides

closed in to prevent radiation of rf energy into the environment, and to provide electrical isolation of equipment being tested for susceptibility to EM

fields. Much has been published describing their design and principles of

operation [2-2,2-3,2-4,2-5].

An isometric view, a side view, and a cross section view of a typical TEM

cell are shown in figures 2-1, 2-2, and 2-3, respectively. The cell consists

of a length of rectangular "coaxial" transmission line made of flat metal

plates, with the cell septum acting as the center conductor. The two ends of

2.2
the cell have tapered dimensions to adapt to standard coaxial connectors. The

line and tapered transitions are generally designed to have a nominal characteristic impedance of 50 Q. in order to obtain a low voltage standing wave

ratio (VSWR) in a 50 o. system. RF energy is conducted through the line from a 50 a transmitter connected at the input port to a 50 fi terminating load (Z^)

connected at the output port. The electric field level at the test point in

the cell, midway between the center of the septum and either (outer) wall, is

given by

where E = electric field strength at the test point, V/m,

V = RMS voltage on the septum (center conductor), V,

b = separation distance between the septum and outer walls, m,

P = net power flow through the cell, W, and

Z^ = real part of the cell's characteristic impedance = 50 a.

The TEM or principal mode of operation produces only E- and H-fields that

are transverse (perpendicular) to the direction of energy flow (propagation).

Such waves can carry energy at frequencies down to dc, while the higher order

(waveguide) modes have large attenuation below a so-called cutoff frequency.

A transmission line must have two separate conductors to support a TEM wave.

The wave traveling through the cell has a wave impedance (E/H ratio) of approximately 377 J2, the same as a far-zone plane wave propagating in free

space. Therefore, the H-field is given by

H = E/Z^ = E/377, (2-2)

where H = magnetic field strength at the test point, A/m, and

Zf = wave impedance of free space = 377 fj.

An approximate equation for the characteristic impedance of a rectangular

transmission line is [2-6,2-7]:

Z = 377 (2.3)

^ 4[A.ll„ (sinhiS)] -k

where a, b, and g are shown in figure 2-3 and k is related to the fringing

2.3
capacitance between the edges of the septum and the side walls. For large

gaps (g/a > 0.2) this fringing term is negligible. As an example, consider

the TEM cell used in the Fields Characterization Laboratory at NBS. The

dimensions shown in figure 2-3 are 2a = 1.2 m, 2b = 0.6 m, and w = 0.81 m. In this case, k of eq (2-3) is negligible and the calculated value of 1^ is

50.1 n.

An expression for calculating Z^ of a triplate transmission line with

open sides, in which the upper and lower (grounded) plates have infinite

width, is given by eq (20) in reference [2-8] as

The calculated value of Z^ for din open triplate line having a plate spacing (b

= 0.3 m) and septum width (w = 0.81 m) equal to that of the above NBS TEM cell

is 52.6 Q. Note that this is only slightly higher than the 50 n value for the

TEM cell with closed sides.

The upper useful frequency of a TEM cell is limited by field distortions

which are caused by waveguide-mode resonances that occur at frequencies above

the empty cell's multimode cutoff. Because of the tapered section at either

end of the cell, the resonant length is not well defined. As a first approximation, the overall cell length is often considered to be the wavelength of

the lowest frequency resonance [2-2]. In other words, for good operation the

spacing between the cell plates, the width of the septum, and the length of

the nontapered section should all be less than x/2. Thus it is generally not

possible to use a TEM cell for calibrating or taking data with most EMI

antennas.

The estimated measurement uncertainty in establishing EM fields in a TEM

cell operating in the fundamental (TEM) mode ranges from ±0.5 dB to ±2 dB,

depending on the shape of the cell (width to height ratio) and the quality of

design. Details for calculating the electric field distribution inside a TEM

cell and procedures for performing radiated susceptibility tests are given

elsewhere [2-6,2-9,2-10].

It is important that the size of the device to be tested does not exceed

approximately one-third of the plate spacing of the cell [2-11,2-12]. This is

because the device tends to short out the test field in the region it occupies

2.4
between the plates, increasing the field amplitude. Field strength errors and

perturbations caused by placing equipment inside the cell can be partially

corrected by measuring the field in the test region near the DUT with a small

calibrated E-field probe, and making appropriate corrections. This procedure

is discussed in reference [2-9].

2.3 Parallel-Plate Line with Open Sides and 377 « E/H Ratio

One major limitation of a TEM cell is the inverse proportionality between

its physical size and upper useful frequency. As mentioned, the spacing between the cell conductors, the width of the septum, and length of the uniform

(nontapered) line section should all be less than x/2. Thus, when designing

an rf chamber for exposing relatively large samples, rather than for calibrating a small measurement probe, it is often not possible or convenient to use a TEM cell. Another disadvantage is the large power required for a given

field intensity, if the characteristic impedance and load resistance are only

50 a. The required power, however, is about 10-20 dB less than that required

for radiated field methods performed in an anechoic chamber or at an open

field site.

Any sharp bends or inhomogenities in a transmission line tend to promote

higher-order waveguide modes. For example, the side walls of a TEM cell perpendicular to the septum permit higher-order modes to propagate at lower frequencies than in a parallel-plate line having open sides [2-13]. Also, it is

sometimes advantageous for other reasons to use open parallel plates or a transmission line having a characteristic impedance other than 50 a, as discussed in section 2.4.

If a parallel-plate transmission line is terminated in its characteristic

impedance, the E/H ratio has a free-space value of 377 a. Also, similar to a plane wave in free space, the E, H and propagation vectors are all mutually

orthogonal. That is, the E and H vectors are perpendicular to each other and

both are perpendicular to the propagation vector of the traveling wave.

Figure 2-4 is a sketch of a parallel-plate transmission line (stripline)

which is useful for generating known E- and H-fields at frequencies up to 30 or 40 MHz. This type of calibrating line is specified in MIL-STD-462 [2-14]

and SAE Standard AIR-1209 as a test chamber for radiated susceptibility (RS)

measurements on small equipment at frequencies between 14 kHz and 30 MHz.

2.5
Notice 3 of the MIL-STD permits the installation of a "long-wire" antenna in a screenroom, for RS testing of equipment that is too large for the stripline

(see chapter 3 of this report).

The expression for calculating the E- and H-field intensity existing at

the test point midway between two flat conducting plates of a stripline is the same as that for a TEM cell, which is given by eqs (2-1) and (2-2). In this

case,

E = electric field strength midway between the two plates, V/m,

V = RMS voltage difference between the plates, V, and

b = plate spacing, m.

Evaluation of a typical MIL-STD-461/462 parallel plate line at NBS was accomplished in three phases, as follows:

(a) measurement of the characteristic impedance and input impedance

to the stripline for various values of terminating resistance and for

various locations of the stripline inside the NBS screenroom, at frequencies from 1 to 100 MHz;

(b) calibration of a small transfer probe suitable for checking the

E field generated within the stripline, and

(c) measurement of the absolute level and uniformity of the generated field strength at frequencies from 2 to 100 MHz using the above

calibrated probe.

2.3.1 Characteristic Impedance and Input Impedance of a Stripline

The general expression for the characteristic impedance of a transmission

line is given by eq (2-5), where the approximation given applies to a low loss

line,

where R = resistance of the line per unit length, fi/m,

G = conductance of the line per unit length, S/m,

2.6
L = inductance of the line per unit length, H/m, and

C = capacitance of the line per unit length, F/m.

The required value of load impedance for terminating a parallel -pi ate

line in its characteristic impedance (Z^-), which results in a minimum voltage

standing wave ratio (VSWR), depends mainly on the plate spacing and width.

The Z^ value of an open-sided line is also affected by its surroundings,

especially if the line is located in a shielded enclosure (high Q cavity) such

as a screen room. Therefore, the optimum value of terminating resistor for a parallel-plate line should be verified experimentally from tests of the Efield uniformity along the line, with a small dipole probe, as discussed in

section 2.3.2 [2-15]. The parallel-plate transmission line of figure 2-4, as fabricated according to MIL-STD-462, has measured values of characteristic

impedance which range from 80 to 160 n, as will be shown later. Similar

striplines with smaller dimensions are also in use. These are useful at frequencies up to 50 or 100 MHz, depending on size.

MIL-STD-462 specifies the use of a stripline having a plate spacing of

0.457 m (1.5 ft) and an upper plate width of 0.610 m (2 ft), as shown in

figure 2-5. This stripline is used for making radiated susceptibility (RS04)

tests of electronic equipment in a known EM field. The width of the lower

plate in figure 2-5 is 0.305 m (1 ft) greater, on one side, than the upper

plate. The lower plate thus has a total width of 0.914 m (3 ft). The MIL-STD

and the SAE documents state that the expected Z^ value for this line is

83 a. Several equations are given in the literature for calculating the

characteristic impedance of parallel-plate lines. For example. Sensor and

Simulation Note XXI by Baum [2-8] gives an equation for Z^ of a balanced

parallel-plate line or a single-plate line above an infinite ground plane.

The characteristic impedance of a two-plate line in free space is

h- (3^7) ^^) {1^:^[1^ ^" (^)]}"^. (2-6)

If the extra 0.305 m (1 ft) width in the lower plate of the MIL-STD-462

line is ignored, the calculated value for Zq using eq (2-6) is 162 n. If it

is assumed that the lower plate extends to infinity on each side (infinite

width), the theoretical value of Z^. is half the above value, or 81 a. This is

close to the 83 9. value given by the MIL-STD.

2.7
Volume 4 of White's series of books on EMI/EMC [2-16] contains a graph

(fig. 4.19 on p. 4.34) of the characteristic impedance of parallel-plate

lines. If the lower plate width (w) is equal to or greater than three times

the width of the upper plate, the graph in [2-16] indicates that I^ of the

MIL-STD-462 line would be 113 fi. The theoretical Z^ of the line would thus be

greater than 113 a (rather than 83 a) because the lower plate width is less

than 3 w

Another equation given in the literature for calculating 1^ of a stripline, in which w/b is restricted to a value between 1 and 10, is [2-17]

^ 377

c

" (w/b) + 2.42 - 0.44 (b/w) + (1 - b/w)6-

^^"'^

According to eq (2-7), the theoretical Z^ value for a MIL-STD-461 line,

assuming infinite width in the lower plate, is 110 ^, Thus the actual MIL-STD

line should have a Z^ nearly double this value. It can be seen that the theoretically calculated values of Z^ for a stripline having an upper plate width

of 0.610 m (2 ft) and a plate spacing of 0.457 m (1.5 ft) range from about 80

to 160 ft, depending on the assumptions used.

A parallel-plate transmission line was fabricated at NBS according to the

specifications and dimensions given in MIL-STD-462. Measurements were done to

determine experimentally the optimum value of terminating load resistor R^ for

this line to see how closely this compares with the specified value of 83 a. These measurements were performed with an automatic network analyzer, which

determines Z^ from measurements of the line input reactance as a function of

signal frequency, for two cases. In one case the line was terminated with a good short circuit and in the other case with an open circuit. That is,

Z = /TTC", (2-8)

c r ^ '

where Z^ = characteristic impedance of the transmission line, a,

Lq = short-circuit inductance, H, and

Cq = open-circuit capacitance, F.

In measuring the line inductance, two short copper straps were soldered

between the tapered end of the upper plate and the ground plate. The straps were connected in place of the three parallel resistors previously connected

2.8
to terminate the line. Figure 2-6 shows the instrumentation used for mea- suring Z(-, Z^p and the E-field uniformity within the MIL-STD-462 stripline.

Measurements were made of the input impedance of the stripline, with the

line terminated in two different values of resistance (83 n and 120 n). The

stripline was supported on wood sawhorses at a height 1 m above the floor.

These data are given in tables 2-1, 2-2, 2-3 and 2-4 for frequencies ranging

from 1 to 100 MHz. The data show that the measured input impedance varies

with frequency for both values of terminating resistance. The optimum value

of terminating resistance appears to be in the neighborhood of 120 q rather

than the 83 a specified in MIL-STD-462. Graphs of the data from tables 2-3

and 2-4 for frequencies up to 50 MHz are shown in figure 2-7.

The measured values of line characteristic impedance are given in

table 2-5 for three different locations of the line inside the NBS screenroom.

Position A corresponds to placement of the stripline near and parallel to the

long wall of the room at a height 1 m above the floor. A copper grounding

strap 10 cm wide was connected in some of the tests between the grounded plate

of the line and the copper-clad table. Position B is the same as position A

except that the stripline was placed on the floor so that the grounded side of

the line was capacitivly coupled to the grounded floor of the screenroom.

Position C refers to the stripline mounted on sawhorses along the center line

of the screenroom, with the grounded plate of the line at a height 1 m above

the floor. Position D is the same as position C except that the stripline was lowered to the floor level. Table 2-5 shows that the measured characteristic

impedance of the transmission line varies greatly with frequency, but for the

lower frequencies it has a value of about 100 q. Graphs of the measured line

Z(- for frequencies up to 50 MHz are shown in figure 2-8 for three positions of

the stripline in the screenroom.

2.3.2 Field Level and Uniformity in a Stripline

The main criterion for a good standard-field setup is the uniformity of

the generated field as a function of frequency. It is also important to know

how closely the test field approximates a uniform plane wave with a wave impedance (E/H ratio) equal to that of free space, which is 377 q. Measurements were made with a calibrated NBS E-field probe (EFM-5) having dipole

antennas 5 cm in length [2-18]. A block diagram of the setup used is given in

2.9
figure 2-6. The field level chosen for these tests was about 5 V/m. The

tests were done to compare the actual measured field with the values

calculated from eq (2-1) given previously, E = V/b.

For most of the tests, the EFM-5 probe was placed midway (vertically)

between the two plates and in the center (horizontally) of the line, as indicated in figure 2-6. The electric field uniformity was checked as a function

of signal frequency using two different values of terminating resistance (83

and 100 n). The value of R in figure 2-6, used to terminate the coaxial cable

leading to the stripline in 50 n, was 145 a when the stripline termination was

83 u. The measured dc input resistance for this case was 54 a. The value of

R was changed to 100 a when the stripline was terminated in 100 fl, and the

measured dc input resistance for this case was 51 a. Measurements of electric

field strength at the probe position, versus signal frequency, were made for

two different locations of the stripline in the empty screenroom. The large

walk-in doors to the room were closed when taking field strength data.

The measured values of electric field strength at the center of the

stripline are shown in table 2-6 and figures 2-9 and 2-10 for both values of

terminating resistance. It can be seen that the standard field values calculated from eq (2-1) are correct within ±2 dB at all frequencies up to

40 MHz. The measured values of E-field at 30 MHz as a function of position

along the line are given in table 2-7 and figures 2-11 and 2-12. It can be seen that the field level is correct within 2.3 dB and uniform within ±0.5 dB over the 2 m length of line at this frequency. The use of a broadband step-up

transformer at the stripline input would result in higher E-field magnitudes

for a given amount of rf power than that obtained with the instrumentation of

figure 2-6.

2.3.3 Effect of Screenroom Resonances on Field Strength

The characteristic impedance of a stripline with open sides is influenced

by the proximity of nearby metal surfaces and the Q of the enclosure

(screenroom) in which it is located. The effect of room resonances on the

behavior of the stripline is clearly evident by the excursions in impedance

and measured field strength at frequencies above 40 MHz. These resonance

effects would occur at lower frequencies if a larger screenroom were used.

The NBS screenroom was initially evaluated for EM shielding effectiveness in

2.10
the frequency range of 10 kHz to 1000 MHz, using MIL-STD-285 as a guide.

However, the data are not included in this report.

Test methods RE02 and RS02 in MIL-STD-462 require that E- and H-field

tests be performed on electronic equipment for radiated emissions and susceptibility. It is generally necessary to make these measurements inside a screenroom. Because of reflections from conducting surfaces, it is difficult

to make measurements that correlate with open-site data. At higher frequencies there are field strength anomalies due to cavity-mode resonances of the

room. The resonant frequencies of an empty rectangular screenroom are given

by

F = 150

/(f)'

- (^ ^ i^)\ (2-9)

where F = frequency, MHz,

a,b,c = cavity mode integers, only one of which can be zero, and

)l,w,h = length, width, height of the screenroom, m, and ji > w > h.

For the NBS screenroom, i = 7.202 m, w = 3.362 m, and h = 2.805 m. In this case, the lowest-frequency cavity-mode resonance is the 110 mode, v;here a

= 1, b = 1, and c = 0. The calculated frequency of this mode is

'

- 15° /^yrk)' ' (3362^' - '''' TM^- (2-10)

The next higher-frequency mode is the 101 mode, which has a calculated frequency of 57.39 MHz. Table 2-8 gives a list of all the possible cavity-mode

resonances for mode integers up to 4. The lowest resonance at 49.24 MHz is

associated with a transverse electric (TE) wave traveling in the i direction.

2.4 Operation of a Parallel -PI ate Line in the Open-Circuit Mode

or Short-Circuit Mode

A parallel-plate stripline (or a TEM cell) can be operated without a resistive termination for frequencies up to 30 or 40 MHz. When the line is

open circuited it acts as a parallel-plate capacitor rather than a transmission line, and produces a field with a high wave impedance (high E/H

ratio). High-level E-fields can then be produced by connecting a broadband

step-up transformer at the input terminal of the line.

2.11
Figure 2-13 is a sketch of a system using open-ended parallel plates to

generate a known electric field. The expression for calculating the E-field

intensity at the point P midway between the flat metal plates is the same as that for a stripline or TEM cell, namely eq (2-1). The uncertainty of the

calculated field is also about the same. The wave impedance, however, is

greater than 377 n. A high level E-field can thus be produced with a low

power source, especially if the line is resonated by means of an inductor, as shown in the figure. For example, a field level of 1000 V/m can generally be

achieved at the resonant frequency with a 1 W power source. At higher

frequencies (above 20 or 30 MHz) approaching self resonance of the system,

similar to a MIL-STD-462 stripline, it is not possible to calculate the field

level from eq (2-1). It is then necessary to measure the field strength at

the calibrating point with a small transfer probe that has been calibrated

previously in a small stripline or a TEM cell, as explained in section 2.5 of

this report. By contrast, when a MIL-STD-462 stripline (or a TEM cell) is

short circuited, it acts as a single-turn loop and produces a field having a

"^ery low wave impedance. This configuration is useful for producing a highlevel H-field with a low E/H ratio.

2.5 Use of E- and H-Field Probes as Transfer Standards

The term "transfer standard" refers to an electrically small antenna (rf

probe). This can be a short dipole for sensing E-fields or a small loop for

H-fields that has a known response over a given range of frequency and amplitude. Such a probe can be used to measure the field strength as a function of

frequency in a transmission line. A typical calibration curve for a 5 cm dipole probe (EFM-5) is shown in figure 2-14.

As mentioned in section 2.3.2, tests of field level vs frequency and

position in a MIL-STD-462 stripline can be made with this type of probe. Such

transfer probes can also be used to improve the construction and operation of

standard-field chambers, for example, by optimizing the value of terminating

load impedance. The optimum value should be determined experimentally for

each measurement situation and location of the line with respect to its sur- rounding objects, screenroom walls, etc.

Transmission line chambers can often be used at frequencies above those

for which the theoretical equations apply. In this case the field strength is

2.12
determined or checked with a transfer probe that has been calibrated previously in a smaller chamber, or which has been calibrated at a lower frequency but is known to have a flat response up to the frequency of interest.

For example, the E- or H-field in a screenroom that has been converted into a long-wire chamber can usually be determined more accurately with a transfer

probe than by relying on theoretical equations.

2,6 References

[2-1] Kummer, W. H. ; Gillespie, E. S, Antenna measurements--1978. Proc.

IEEE. 66(4): 483-507; 1978 April.

[2-2] Ma, M. T.; Kanda, M.; Crawford, M. L. ; Larsen, E. B. A review of

electromagnetic compatibility/interference measurement methodologies,

Proc. IEEE. 73(3): 388-411; 1985 March.

[2-3] Crawford, M. L. Generation of standard EM fields using TEM

transmission cells. IEEE Trans, on EMC. EMC-16(4): 189-195; 1974

November.

[2-4] Crawford, M. L. Generation of standard EM fields for calibration of

power density meters 20 kHz to 1000 MHz. Nat. Bur. Stand. (U,S,) NBSIR

75-804; 1975 January,

[2-5] Mitchell, J, C, A radio frequency radiation exposure apparatus.

Brooks Air Force Base, School of Aerospace Medicine, Technical Report

SAM-TR-70-43; 1970 July,

[2-6] Tippet, J. C; Chang, U,C, Radiation characteristics of dipole sources

located inside a rectangular coaxial transmission line, Nat, Bur,

Stand, (U.S.) NBSIR 75-829; 1976 January.

[2-7] Chen, T. S. Determination of the capacitance, inductance, and

characteristic impedance of rectangular lines. IRE Trans, on MTT.

MTT-8(5): 510-519;; 1960 September.

[2-8] Baum, C. E. Note XXI. Impedances and field distributions for parallel

plate transmission line simulators. Electromagnetic Pulse Sensor and

Simulation Notes. Vol. 1, DASA Report 1880-1; 1967 April.

[2-9] Crawford, M. L.; Workman, J. L, Using a TEM cell for EMC measurements

of electronic equipment, Nat, Bur, Stand, (U.S,) Tech, Note 1013;

Revised 1981 July, 65 p,

[2-10] Crawford, M, L, Improving the repeatability of EM susceptibility

measurements of electronic components when using TEM cells.

International Congress Exposition; 1983 February 28-March 4; Detroit,

MI; and SAE Handbook; 1984.

2,13
[2-11] Kanda, M. Electromagnetic-field distortion due to a conducting

rectangular cylinder in a transverse electromagnetic cell. IEEE Trans,

on EMC. EMC-24(3): 294-301; 1982 August.

[2-12] Kanda, M. Analytical and numerical techniques for analyzing an electrically short dipole with a nonlinear load. IEEE Trans. Ant. and

Prop. AP-28: 71-78; 1980 January.

[2-13] Weil, C. M. ; Joines, W. T.; Kinn, J. F. Frequency range of large-scale

TEM mode rectangular strip lines. Microwave Journal: 93-100; 1981

November.

[2-14] Military Standard, Electromagnetic Interference Characteristics,

Measurement of. MIL-STD-462; 1967 July 31.

[2-15] Roseberry, B. E.; Schulz, R. B. A parallel-strip line for testing RF

susceptibility. IEEE Trans, on EMC. EMC-7: 142-150; 1965 June.

[2-16] White, D. R. J. EMC Handbook, Vol. 4, EMI test instrumentation and

systems. Maryland: Don White Consultants, Inc.; 1971.

[2-17] Gardiol, F. E.; Zurcher, J. F. HP-65 program computes microstrip

impedance. Microwaves: 186; 1977 December.

[2-18] Larsen, E. B.; Ries, F. X. Design and calibration of the NBS isotropic

electric-field monitor (EFM-5), 0.2 to 1000 MHz. Nat. Bur. Stand.

(U.S.) Tech. Note 1033; 1981 March. 103 p.

2.14
Table 2-1. Measured input impedance of the NBS MIL-STD-462 stripline

in an unshielded laboratory, with 83 fj termination.

Frequency

MHz

Resistance Reactance Impedance

Q. Ratio

Zin/83

1 83 5 83 1.00

1.5 83 7 83 1.00

2 83 10 84 1.01

2.5 84 12 85 1.02

3 85 15 86 1.04

3.5 87 15 88 1.06

4 84 18 86 1.04

4.5 84 25 88 1.06

5 88 26 92 1.11

5.5 91 28 95 1.14

6 89 28 93 1.12

6.5 93 36 100 1.20

7 98 38 105 1.27

7.5 101 39 108 1.30

8 105 41 113 1.36

8.5 107 41 115 1.39

9 111 43 119 1.43

9.5 116 43 124 1.49

10 121 42 128 1.54

15 157 22 159 1.92

20 147 -32 150 1.81

25 111 -26 114 1.37

30 97 -9 97 1.17

35 93 -1 93 1.12

40 86 11 87 1.05

45 114 28 117 1.41

50 119 10 119 1.43

55 105 19 107 1.29

60 108 40 115 1.39

65 168 59 178 2.14

70 132 -28 135 1.63

75 118 -14 119 1.43

80 97 10 98 1.18

85 93 29 97 1.17

90 135 59 147 1.77

95 147 16 148 1.78

100 127 12 128 1.54

2.15
Table 2-2. Measured input imp»edance of the NBS MIL-STD-462 strip line

in an unshielded 1 aboratory, with 120 fl termination.

Frequency Resistance Reactance Impedance Ratio

MHz Q Q. Q. Zin/120

1 120 -2 120 1.00

2 118 -3 118 0.98

3 116 -7 116 0.97

4 108 -3 108 0.90

5 115 -2 115 0.96

6 108 1 108 0.90

7 114 1 114 0.95

8 115 115 0.96

9 115 115 0.96

10 115 -2 115 0.96

11 113 -2 113 0.94

12 112 -1 112 0.93

13 109 -3 109 0.91

14 110 5 110 0.92

15 113 4 113 0.94

16 114 4 114 0.95

17 115 4 115 0.96

18 118 3 118 0.98

19 118 -1 118 0.98

20 116 -4 116 0.97

21 111 -8 111 0.93

22 102 -1 102 0.85

23 103 8 103 0.86

24 108 12 109 0.91

25 110 13 111 0.93

26 114 16 115 0.96

27 120 16 121 1.01

28 124 13 125 1.04

29 125 12 126 1.05

30 126 11 126 1.05

35 126 -21 128 1.07

40 91 -28 95 0.79

45 89 5 89 0.74

50 100 15 101 0.84

55 93 15 94 0.78

60 97 45 107 0.89

65 128 71 146 1.22

70 203 -19 204 1.70

75 120 -38 126 1.05

80 93 -8 93 0.78

85 89 19 91 0.76

90 110 54 123 0.92

95 137 28 140 1.17

100 135 37 140 1.17

2.16
Table 2-3. Measured input impedance of the NBS MIL-STD-462 stripline

in center of NBS screenroom, with 83 fl termination.

Frequency

MHz

Resistance Reactance Impedance Ratio

Zin/83

1 83 4 83 1.00

1.5 83 6 83 1.00

2 83 9 83 1.00

2.5 84 12 85 1.02

3 85 15 86 1.04

3.5 87 15 88 1.06

4 84 18 86 1.04

4.5 84 25 88 1.06

5 88 26 92 1.11

5.5 91 28 95 1.14

6 89 28 93 1.12

6.5 93 36 100 1.20

7 98 38 105 1.27

7.5 101 39 108 1.30

8 105 41 113 1.36

8.5 107 41 115 1.39

9 111 43 119 1.43

9.5 116 43 124 1.49

10 121 42 128 1.54

15 157 22 159 1.92

20 147 -32 150 1.81

25 111 -26 114 1.37

30 97 -9 97 1.17

35 93 -1 93 1.12

40 86 11 87 1.05

45 114 28 117 1.41

50 119 10 119 1.43

55 105 19 107 1.29

60 108 40 115 1.39

65 168 59 178 2.14

70 132 -28 135 1.63

75 118 -14 119 1.43

80 97 10 98 1.18

85 93 29 97 1.17

90 135 59 147 1.77

95 147 16 148 1.78

100 127 12 128 1.54

2.17
Table 2-4. Measured in piut impedance of the NBS MIL-STD-462 stripl ine

in center of NBS screenroom. with 120 fi termination.

Frequency Resistance Reactance Impedance Ratio

MHz a Q -

Q. Zin/120

1 120 -2 120 1.00

2 118 -3 118 0.98

3 116 -7 116 0.97

4 108 -3 108 0.90

5 115 -2 115 0.96

6 108 1 108 0.90

7 114 1 114 0.95

8 115 115 0.96

9 115 115 0.96

10 115 -2 115 0.96

11 113 -2 113 0.94

12 112 -1 112 0.93

13 109 -3 109 0.91

14 110 5 110 0.92

15 113 4 113 0.94

16 114 4 114 0.95

17 115 4 115 0.96

18 118 3 118 0.98

19 118 -1 118 0.98

20 116 -4 116 0.97

21 111 -8 111 0.93

22 102 -1 102 0.85

23 103 8 103 0.86

24 108 12 109 0.91

25 110 13 111 0.93

26 114 16 115 0.96

27 120 16 121 1.01

28 124 13 125 1.04

29 125 12 126 1.05

30 127 10 127 1.06

35 126 -21 128 1.07

40 91 -28 95 0.79

45 89 5 89 0.74

50 100 15 101 0.84

55 93 15 94 0.78

60 97 45 107 0.89

65 128 71 146 1.22

70 203 -19 204 1.70

75 120 -38 126 1.05

80 93 -8 93 0.78

85 89 19 91 0.76

90 110 54 123 1.03

95 137 28 140 1.17

100 135 37 140 1.17

2.18
Table 2-5. Measured characteristic impedance of MIL -STD-4 62

stripline in NBS screenroom.

Ch<jracteristic Impedance, ohms

Frequency,

MHz Position

102

A Position C

111

Po sition D

2 110

4 101 132 108

6 100 122 107

8 105 117 105

10 142 121 105

12 100 119 102

14 88 112 99

16 63 94 57

18 392 312 155

20 135 143 126

22 120 129 118

24 113 134 114

26 106 136 108

28 99 134 102

30 86 135 93

32 40 123 50

34 154 151 423

36 209 103 145

38 148 169 130

40 136 143 125

42 125 139 118

44 113 138 106

46 68 132 83

48 279 836 375

50 74 41 76

55 91 51 65

60 362 115 325

65 138 124 347

70 316 131 216

75 137 217 117

80 99 78 807

85 55 74 78

90 143 305 77

95 295 193 64

100 159 168 196

2.19
Table 2-6. Measured E-field error in center of MIL-STD-462 stripline

Stripline error with

83 Q termination, dB

Stripline error with

100 Q termination, dB

Frequen

MHz

cy

Position A*

0.0

Position R

0.6

Position A*

0.2

Position A**

0.8

Position B

2 0.7

4 - 0.6 0.6 1.3 0.8

6 0.0 0.5 0.3 1.8 0.9

8 - 0.3 0.7 1.1 2.8

^10 0.0 0.5 0.8 1.6 1.2

12 - 0.2 0.8 0.8 1.4

14 0.0 0.6 0.8 1.3 1.2

16 - 0.3 0.1 -0.1 1.3

18 0.1 0.8 0.7 0.7 1.2

20 - 1.0 0.9 1.1 1.3

22 0.8 1.5 0.8 1.1 1.6

24 - 0.4 1.4 0.7 -0.2

26 1.8 1.4 0.9 1.6 -0.1

28 - 1.6 1.2 1.5 -0.1

30 2.6 2.1 0.9 1.4 -0.1

32 - 1.4 1.0 1.2 -0.4

34 2.1 1.5 0.5 0.8 -0.8

36 - 1.0 0.1 0.1 -0.9

38 1.2 0.6 0.2 -0.2 -1.2

40 - -0.2 -0.8 -0.7 -1.7

42 1.6 -0.2 1.8 -1.2 -1.2

44 - -1.1 -2.0 -1.0 -1.6

46 -4.5 -2.4 -4.2 -3.7 -2.8

48 -10.5 -6.1 -12.3 -12.4 -12.2

50 9.1 -3.7 8.7 8.7 3.7

52 - 1.7 5.6 5.8 1.8

54 3.0 1.1 3.6 3.9 0.9

56 - 0.0 3.1 3.3 -0.8

58 - 0.5 2.3 2.8 0.0

60 3.5 0.4 2.5 2.2 1.3

*No ground strap

**With strc'P

2.20
Table 2-7. Measured E-field error along center line of MIL-STO-462

stripline, at 30 MHz

Stripline error with

83 Q termination, dB

Stripline error with

100 Q termination, dB

Distance from

center of

stripline, cm Position A* Position B Position A* Position A** Position B

-90 2.6 2.0 0.9 0.9 -0.2

-75 2.8 2.1 1.0 1.0 -0.2

-60 2.8 2.1 1.0 1.0 -0.2

-45 2.9 2.2 1.0 1.0 -0.2

-30 2.9 2.2 1.0 1.0 -0.2

-15 2.9 2.3 1.0 1.1 -0.2

2.9 2.3 1.0 1.1 -0.1

15 2.8 2.2 1.0 1.1 -0.1

30 2.7 2.1 1.0 1.1 0.0

45 2.6 2.1 1.1 1.1 0.1

60 2.4 2.0 1.1 1.1 0.1

75 2.2 1.9 1.1 1.1 0.2

90 2.0 1.7 1.1 1.1 0.3

105 1.8 1.6 1.1 1.0 0.3

*Mo ground strap

**With strap

2.21
Table 2-8. Calcu lated cavity-m(

first four modes.

Resonance Mode Frequency,

Number Integers MHz

1 1 1 49.24

2 1 1 57.39

3 2 1 61.04

4 2 1 67.79

5 1 1 69.64

6 1 1 1 72.69

7 3 1 76.78

8 2 1 1 81.15

9 3 1 82.24

10 1 2 91.63

11 3 1 1 93.56

12 4 1 94.51

13 2 2 98.48

14 4 1 99.0

15 2 1 104.03

16 1 2 1 106.09

17 4 1 1 108.59

18 3 2 108.93

19 1 2 108.96

20 2 2 1 112.06

21 2 2 114.78

22 1 2 115.88

23 1 1 2 117.74

24 3 2 1 121.35

25 4 2 122.08

26 2 1 2 123.14

27 3 2 123.87

28 3 1 2 131.66

29 4 2 1 133.28

30 1 3 135.46

31 4 2 135.57

32 2 2 139.29

33 2 3 140.18

34 1 2 2 140.84

35 4 1 2 142.72

36 3 1 144.14

37 2 2 2 145.38

38 1 3 1 145.63

39 3 3 147.71

40 2 3 1 150.03

41 3 2 2 152.66

42 3 3 1 157.1

43 4 3 157.66

44 1 3 161.77

45 4 2 2 162.3

46 2 3 165.75

47 4 3 1 166.48

48 1 3 166.52

49 1 1 3 167.81

50 3 2 171.33

51 2 1 3 171.65

52 3 3 172.17

53 1 3 2 172.59

54 2 3 2 176.32

55 3 1 3 177.85

56 1 4 179.68

Resonance Mode Frequency

Number Integers MHz

57 4 3 180.77

58 3 3 2 182.37

59 2 4 183.26

60 2 3 183.57

61 1 2 3 184.75

62 4 1 3 186.19

63 4 1 186.3

64 1 4 1 187.47

65 2 2 3 188.24

66 3 4 189.09

67 4 3 2 190.51

68 2 4 1 190.9

69 3 2 3 193.92

70 3 4 1 196.5

71 4 4 196.95

72 4 2 3 201.59

73 4 4 1 204.08

74 4 2 208.06

75 3 3 208.93

76 1 4 2 209.1

77 1 3 3 209.97

78 2 4 2 212.19

79 2 3 3 213.04

80 1 4 214.92

81 3 4 2 217.24

82 2 4 217.92

83 3 3 218.07

84 1 218.51

85 1 1 219.5

86 2 1 222.44

87 3 222.84

88 4 4 224.12

89 4 3 224.93

90 3 1 227.27

91 4 229.55

92 2 231.77

93 1 2 232.7

94 4 1 233.85

95 2 2 235.48

96 4 239.97

97 3 2 240.04

98 1 4 240.87

99 2 4 243.56

100 4 2 246.29

101 3 4 247.97

102 3 252.33

103 1 3 253.19

104 4 4 254.02

105 2 3 255.74

106 3 3 259.95

107 4 3 265.73

108 4 278.58

109 1 4 279.35

110 2 4 281.67

111 3 4 285.5

112 4 4 290.77

2.22
ot-«a •oc

<u

4-> ID

t-QiC

<uo

"O

0)

(/) 3

<U

u

o.

c (0

(DE

»S2 Q.

3 XJ cr c

UJ z>

^o •

>!-> IT) Q) -O ^ r—

I/) 0)

•p- O «4- •»-s_ n:

•-> a> -a E c

o lO w

CLlX. c ^^

I CM

0)

cn

2.23
TEM CELL (Z„ « 5012, E/H s 377n)

CENTER CONDUCTOR (SEPTUM)

b PROBE

Figure 2-2. Side view of a typical TEM cell.

2b —g— -*— g—

2a ^

Figure 2-3. Cross-sectional view of a TEM cell

2.24
Figure 2-4. Isometric sketch of a typical MIL-STD-462 parallel -pi ate

transmission line.

K w • 0.610 m (2 ft) >1

Field calibrating point^^

k-0.305 m -

I (1 ft) •

b = 0.457 m (1.5 ft)

- ±

Figure 2-5. Cross-sectional view of the NBS MIL-STD-462 parallel-plate

transmission line.

2.25
a'=«o

r"

kX o1— ^ tn

LU oLUO

=>ooc

CD 1 5 ^^

0)c OO => ^ """^ •r- CO DC y V

LOW

PA

FILTE

OBE

-FIELD PROBE EFM-5) >

r— l/ • 1 ^ S >^

i. LU *-^ "O LU ^ >^«^ \ p*M I'MK ^ 0) H^ & s n—

i. \ y^ ^ tu

«- / \ I 1 LU

ex: \ J 1—

t- i - «« OJ to ^ V _J .C : ^ +*

oc o> ^ LU c 1 ^: 3 •^

ex: cc

p

'; • O ^ -J 3

<C LU tn •^M

LU i-H 1 iO Z Ll. d ^

:s •—1 1— i t j V -J -J

Ll- s:

s: ; *: l_ !> S o

QC «c LU ^ H-

_l

fiO ^ r-

^* ll. 0£ |i co • o -1 LU

1— -r- Q)

4->

C 5o

LU

n *0 •!- / \ J

4J .— o —1 o Uz c: t—* > =5 *J o L 1- W ^ 4^ in / V) cvj ' c vo

•-na- cc si° OH • z^ < >A/V^ — !• 1 CM ^ UJ (U C3 \, =3 If u.

2.26
175

150

U)£

I

O 125

• u

o

z

100

cca

u

Q.£ 75 M

H

D

Q.Z 50

(n£

I

O

u

u

z

ccQU

£

25

63 OHM TERMINATION +

120 OHM TERMINATION x

10 20

FREQUENCY^ MHz

30 40 50

Figure 2-7. Measured input impedance of the MIL-STD-462 stripline

in the center of the NBS screenroom. 800

700

600

500

400

uM

M 300

u

(C 200

q:a:z

" 100

T -r -r POSITION R X

POSITION C + POSITION D o

20

FREQUENCY, MHz

Figure 2-8. Measured characteristic impedance of the MIL-STD-462

stripline in the NBS screenroom.

2.27
10

8

m

•D 'd

_ PI q:o

q: -2

u

a -4

_i

uM -B li.

1 U -B

-10

-12

-14

63 OHM TERMINRTION

100 OHM TERMINRTION

I I I I I I 10 15 20 25 30 35 40 45 50

FREQUENCY, MHz

Figure 2-9. Measured E- field error in center of stripline.

Position A in screenroom, with no ground strap

10 T

B

CD

•o

q:o

q:

q:u

Q

uM

I u

-2

T

-4|-

-G

-8

-10

-12

-14

63 OHM TERMINRTION

100 OHM TERMINRTION

J. I 10 15 20 25 30 35 40

FREQUENCY , MHz

Figure 2-10. Measured E- field error in center of stripline.

Position B in screenroom.

45 50

2.28
2 - o

O

q:u 1 Q

J

u

b.

I u

-1

I I

-90

T r

t T r T p

N w w w w

83 OHM TERMINATION +

108 OHM TERMINATION x

J L. ' ' J L I

I 1

-w w I

J L. -60 -30 30 60 90

DISTANCE FROM CENTER. Cm

Figure 2-11. Measured E-field error along length of strlpline at 30 MHz,

83 a termination. Position A in screenroom, with no groundstrap.

T 1 r I I -r r I I

m

o

q:

(^ 1 Q

-IuM

It.

1bJ

-1

J L X

83 OHM TERMINATION +

100 OHM TERMINATION x

I JL -90

Figure 2-12

-60 -30 30

DISTANCE FROM CENTER, Cm

60 90

Measured E-field error along length of strlpline at

30 MHz, 100 Q termination. Position B in screenroom. 2.29
SIGNAL

GENERATOR

-» RF LINEAR

AMPLIFIER

LOW PASS

FILTER

50 Q COAX CABLE

RESONATING

COIL

1 m

0.5 to 1 m

Added capacitance

at low frequencies (1) Top and bottom of cage are pieces of metal -sided olywood.

(2) Sides of cage are 2x4

wood supports.

, (3) P » Optimum field point.

Figure 2-13. A parallel -pi ate cage system for generating a known E-field,

0.01 to 40 MHz.

RF

VOLTMETER

>

CQ

a

u.D

(D

(0 _o

c

,X) 1 mn"1 1 1 1 iiiii| 1 1 1 1 iiii| "T"'1 1 iiiiij 1 1 1 34 - A

-

32 - /I

"

30 -

/

1

28 -

f 1 -

26 - 1

24 - 1-

22

20

18

_ _^^

lr 1-

16 - l14 - l

12 1

1 1 Mini 1 1 1 1 mil 1 1 1 1 Mill .-_j__1 1 iiiiil 1 1 il 0.1 1 10 100 1000

Frequency, MHz

Figure 2-14. Typical response of an EFM-5 probe vs frequency, at

strength of 20 dBV/m (10 V/m).

5000

a field

2.30
3. EM Field of a Single-Wire Transmission Line in a Screenroom

3.1 Introduction

The long-wire antenna is used in a shielded enclosure when test items

are too large to be placed in a parallel-plate line or other standard- field

chamber. This long-wire antenna is illustrated in figure 3.1 installed in a shielded enclosure as per MIL-STD-461/462 specifications. The objective in

testing the long-wire antenna at NBS was to determine the electromagnetic

field uniformity and compare the measured field with theoretically

calculated values

MIL-STD 461/462 describes a laboratory method of simulating, in a shielded room, a radiated field having vertical polarization and

controllable properties of electromagnetic (E and H) field strength. The

long-wire antenna provides a means for applying a known field to test the

susceptibility of electronic equipment at frequencies up to about 30 MHz.

This long-wire antenna, as shown in figure 3-1, is the center conductor of a chamber formed by the walls of the shielded room which act as the outer

conductor. This "coaxial" transmission line is terminated in its

characteristic impedance, so that standing waves are minimized and the field

intensity is fairly constant with respect to frequency and position along

the line.

Two techniques are used for matching the generator impedance to the

characteristic impedance of the long-wire transmission line as follows:

(1) An intermediate impedance -matched line is connected between the longwire input and a short 50 ohm coaxial line from the signal generator, as shown in figure 3-2. If the characteristic impedance of this intermediate

line is R-. and is matched by a parallel combination of R_ and R„ , where R_

is the terminating impedance of the long-wire line (equal to its

characteristic impedance Z_) , the delivered voltage to the long-wire line is

(R^/50)(V ), where the delivered voltage, V , will generate an

3.1
electromagnetic field proportional to the increased voltage. This technique

requires that the 50 ohm short cable from the signal generator be less than

0.025A in length at its highest frequency of operation to minimize standing

waves in the system.

(2) Instead of using a special intermediate transmission line, a 50 ohm

coaxial feed line is used, as shown in figure 3-3. The 50 ohm coaxial line

is terminated with R„ such that the parallel combination of R_ and R„ yields

50 ohms. The electromagnetic field in this case is proportional to V . The

first technique has the advantage of generating more voltage in the longwire antenna, while the second technique has the advantage of simplicity in

construction.

Haber [3-1] describes a method for calibrating loop antennas in a shielded enclosure using the long-wire as its H field generator. His

approach involves generating a known field by an energized wire mounted

near the ceiling of the enclosure. Other formulas for computing the E and H

field magnitudes in the enclosure have been published. For example, Bond [3-

2] considers the system to be equivalent to an infintely long wire under a perfectly conducting plane of infinite extent. Terman and Pettit [3-3] use a similar solution to Bond but include an image antenna below the plane of the

floor.

The premise for using the long-wire antenna is that the field strength

in the chamber is calculable and susceptibility measurements can be made

using the field equations specified by MIL-STD 461/462.

3.2 Experimental Configuration of a Typical Long-Wire Antenna

The long-wire antenna is shown in figure 3-1 as installed in a shielded

room. A No . 10 wire was used in the NBS room, suspended from the two end

walls of the room, driven at one end and terminated at the other. The

resistors, R and R„ remain fixed, once calculated.

The internal dimensions of the NBS shielded enclosure are 7.0 meters

long by 3 . 7 meters wide by 2.4 meters high. These dimensions are minimum

3.2
values for MIL-STD 461/462 emission and susceptibility testing. The (center

conductor) wire was suspended 0.65 meters from the ceiling. With this

geometrical configuration, the characteristic impedance of the long-wire

antenna can be calculated using the equation

Zq=138 log^Q[-^], n, (3-1)

where a=distance of wire to ceiling, m, D=diameter of wire, m.

According to the MIL-STD, this equation for the characteristic

impedance can be used when the distance of the wire to the ceiling is less

than one third the room height. Using equation (3-1), the characteristic

impedance of the long-wire antenna was determined to be 422 Q.

Technique (2), as discussed by White [3-4], was used to determine

experimentally the final values of R^ and R_ for optimum termination of the

long- wire antenna. With the installation of these optimum values for R_ and

R„ , construction of the long-wire is complete. Final experimental

determination of the optimum impedance was 380 Q for R„ and 57 Q for R„

.

3.3 Evaluation of a Long-Wire Antenna

3.3.1 Field Uniformity Along Axis of Long-Wire.

To measure the field uniformity of the long-wire antenna chamber, an EFM-5 probe, developed at the National Bureau of Standards, was used [3-5].

This probe was calibrated in a TEM cell and used as a transfer standard

for measuring the fields of the long-wire antenna [3-6]. The probe has three

mutually orthogonal dipoles to sense the field and can be used to measure

the "total" E field or the three mutually-perpendicular field components

(vertical, horizontal, propagation path). Because of its design and small

size, the EFM-5 does not disturb the field being measured. The EFM-5 was

3.3
used mainly to measure the vertical field component directly beneath the

center conductor of the long-wire.

Figure 3-4 shows the measured field uniformity along the center line of

the screenroom chamber at 1 MHz. Figure 3-5 shows the field uniformity at 15

MHz and figure 3-6 at 30 MHz. The field deviation along the axis of the

chamber was taken with respect to the field measured at the midpoint of the

chamber. Figure 3-4 shows that the deviation was less than ±0.3 dB at 1 MHz,

figure 3-5 shows a deviation less than ± 1 dB at 15 MHz, and figure 3-6

shows a deviation of + to - 3 dB at 30 MHz. The voltage standing wave on

the long-wire antenna at these frequencies is clearly shown.

Figures 3-4, 3-5, and 3-6 indicate that a fairly uniform field can be acheived in the long-wire antenna chamber. However, terminating the

antenna with a resistor equal to its measured characteristic impedance, as specified and described in MIL-STD 461/462, is important. It is also

important that the field uniformity be measured, using a transfer probe that

does not distort the field being measured. The measurement error of the EFM5 is estimated to be less than ± 1/2 dB when calibrated at the measurement

frequencies used in the long-wire antenna screenroom. Using such a transfer

probe standard also has the advantage of checking the absolute value of the

field, as a function of frequency and location in the screenroom.

3.3.2 Measured vs Calculated Field Beneath the Long-Wire

Experimental investigations of the fields in a screenroom, as energized

by a single wire transmission line, indicate that the field uniformity as a function of distance below the long-wire is quite poor. The accuracy of the

calculated electric field intensity below the line depends on careful

optimization of the line terminating impedance. Equations to determine E and

H and the approximate required values of R„ and R„ are given in the

literature [MIL-STD-461 and IEEE Standards on Receivers]. In the derivation

of these equations it is assumed that a pure resistance is an adequate

3.4
termination for the long-wire transmission line. It is also assumed that if

the line height is greater than 2/3 the room height, the presence of the

side walls and floor of the screenroom can be ignored when calculating the

characteristic impedance of the line. The theoretical field strength at a distance d (figure 3-1) below the center of the screenroom long-wire is then

given approximately by:

60 V 1 1 1 1 - (Z^+ Z )^ d ' d + 2a d -2b d -2a -2b

g ,1 1,1 1

"^d +2a +2b " d -2a + 4b "^ d +4a +2b d -4a -4b

and H==E/377

where E=electric field strength at a distance

(3-2)

center of the long-wire, V/m,

H=magnetic field strength at the same point, A/m,

V=RMS voltage input to the transmission line, V,

Z_=characteristic impedance of the line, Q,

Z =impedance of the generator, Q, oa=distance from the wire to the ceiling, m, b=distance from the wire to the floor, m, d=distance from the wire to the field point P, m.

Equation (3-2) was used to calculate the electric field, and this was compared with the measured field (using the EFM-5 transfer standard) for

signal frequencies of 1, 2, 3, 5, 7.5 15, 22 and 30 MHZ. The vertical

component of electric field was used for these comparisons, at several

distances below the center of the long-wire. The differences between

calculated and measured field strength (in dB) were plotted against

distance. This distance ranged from 0.2 meters to 0.8 meters below the

center of the long-wire. The measured field was taken as the dB reference

when compared to the calculated field.

3.5
Figure 3-7 is a plot of this calculated- to-measured field difference

versus distance below the long-wire antenna. It shows that the difference

between the calculated field using eq (3-2) and the measured field varies

both as a function of distance and frequency. The data show that the

calculated E- field has a greater error as the distance from the center

conductor increases

3.4 Conclusions and Recommendations

The data indicate that the average vertical electric field strength

produced in the long-wire antenna chamber is uniform within 3 dB along the

length of the line in the frequency range where the screenroom does not have

any resonances (up to 30 MHz) provided that care has been exercised to

terminate the long-wire antenna in its characteristic impedance (to

eliminate standing waves). However, it is recommended that a transfer

standard probe such as a calibrated EFM-5 be used for measuring the

uniformity of the field and for checking the absolute value of the field.

The error in the calculated field of the long-wire antenna using

equation (3-2) varies with both position and frequency. It is recommended

that a calibrated transfer probe be used to check the absolute value of the

field strength at any desired location in the chamber.

3 . 5 References

[3-1] Haber, F. Generation of Standard Fields in Shielded Enclosures, IRE

Proc. , Vol 42, No. 11 1954.

[3-2] Bond.D.S. Radio Direction Finders, McGraw-Hill Book Co. Inc., New

York, N.Y., pp 221-229 1944.

3.6
3-3] Terman, F.E. and Pettit, J. M. , Electronic Measurements, McGrawHill Book Co. Inc.. New York, N. Y. , pp 405-406 1952.

3-4] White D. Electromagnetic Interference and Compatibility, EMC

Handbook, Vol. 4, pp 4.7-4.13, Don White Consultants 1971.

3-5] Larsen, E.B., and Ries , F.X. Design and Calibration of the NBS

Isotropic Electric-Field Monitor (EFM-5), 0.2 to 1000 MHz, U.S. Nat.

Bur. of Standards, NBS Tech. Note No. 1033, 104 pages March 1981.

3-6] Crawford, M.L. Generation of standard EM fields using TEM

transmisssion cells, IEEE Trans. Electromag. compat. , EMC- 16 No. 4,

pp 189-195 November 1974.

Figure 3-1. The long-wire antenna in a shielded enclosure

3.7
Short 50Q Line * enclosure length ^

Long-Wire Antenna

Figure 3-2. Signal generator drives intermediate feeder line to

long-wire antenna.

• enclosure length > ^0 = ^2

1

Signal

Generator

Long-Wire Antenna

Figure 3-3. Signal generator coaxial line feeds long-wire

directly.

3.8
CD

•D

c

o

CO>

o 01-

o

LU

1 I I I I I 1 I

I I I I I I I II I I I I I I

I

I ; I I I I t

_j,+++'''+++++++++++++++++++++++^

3.2 -1.6 .0 1.6 3.2

Distance from midpoint, m

Figure 3-4. Electric field uniformity of long-wire antenna along

the length of the chamber at 1 MHz

.

CD

•D

C

o

"5*^

(0

>

o

LU -2-

>

I

I I > ' > ' I' —I >

I I I

I

I I I—I

I I—I—I—I—r—i

I I I I

•55 1 - +++++

++++++

++

+++

+++%^^+++++^

'''''''''''''''''''''''''

3.2 -1.6 .0 1.6 3.2

Distance from midpoint, m

Figure 3-5. Electric field uniformity of long-wire antenna along

the length of the chamber at 15 MHz

.

3.9
CO

•D

C

o

.2

CO>

o

o

o

LU

I I I I I I I I

I

I Ill—I I I I—I II I I I I r I I I

.++++

+* +++

: I

, •^^++-^-^1 I,. I ^+++^ ,-

I I I I I TT I I I I I I I 1 I I I I I I I I I I I I I I I I I I I I 3.2 -1.6 1.6 3.2

Distance from midpoint, m

Figure 3-6. Electric field viniformity of long-wire antenna along

the length of the chamber at 30 MHz.

•DO

3 CO

(0 D

m

0) 0)

E c

**o £

o

•D y*

'•B (0

"D 3

O a> (0

•*- O

Distance below center conductor, m

Figure 3-7. Calculated-to-measured field versus distance below

the long-wire for selected frequencies from 1 to 30

MHz.

3.10
4. Calibration of Antenna Factors for EMI Antennas

4.1 Introduction

The measurement of electromagnetic fields in a shielded enclosure

(screenroom) has serious problems because of uncertain antenna factors and

multipath reflections from conductive surfaces. Most electromagnetic

interference antennas at NBS are calibrated in a known field at an open site

or anechoic chamber, using the standard antenna or standard field method.

Because the resulting antenna factors are not necessarily applicable for

making measurements in a screenroom, the errors for measuring field

strength with the calibrated antenna can be quite large. This report

presents the results for antenna factors determined in a screenroom using

the two -antenna method. These antenna factors are compared with antenna

factors determined at an open field site. Experimental data are presented to

show the variability of antenna factor as a function of frequency and

location in the screenroom, thereby providing an indication of error bounds.

The historical background of screenroom measurements is referenced. A

technique is offered for making more accurate field measurements in a screenroom.

The accuracy of measurements in a shielded enclosure has been

investigated in terms of near- field effects [4-1] , theoretical

considerations for making RFI measurements in a shielded enclosure [4-2],

and wall reflections [4-3]. Other errors when making measurements in a shielded room are caused by: antenna VSWR, antenna height above the floor,

dimensions of the enclosure, and antenna detuning by walls and ceilings.

Added to these are instrument errors, and systematic and random errors involved in making the measurement. Because of the many sources of errors,

it is not a simple task to make accurate measurements in a shielded

enclosure. Each of the above variables requires extensive analysis for

improving accuracy. All of these measurement errors indicate that a

4.1
substantially different method from presently established procedures must be

developed.

The following sections will discuss: historical papers dealing with

evaluation of measurements in screenrooms , error bounds of antenna factors

measured in a screenroom, and reliability of measured antenna factors for

making screenroom measurements

.

As background information, a description is given of electromagnetic

(EM) measurements, measurement philosophy, measurement theory, and

measurement errors in a screenroom. For many reasons, a screenroom is often

used to perform electromagnetic radiation measurements, especially at

frequencies below 1 GHz. However, a screenroom produces measurement problems

due to multiple reflections, resonances, and distortion of the

electromagnetic fields. Measurement results are influenced by the size of

the enclosure, location of the test equipment, spacing of the equipment

under test (EUT) and the type of transmitting or receiving equipment.

The frequency and amplitude of screenroom resonances are determined by

the dimensions and geometry of the room, conductivity of the enclosure

surfaces, and by objects inside the enclosure. For an empty rectangular

enclosure the resonant frequencies can be calculated from eq (2-9 of thia

report which is

f=150.

2 2 2 (f) + (-) + (^). (4-1)

where F= frequency, MHz.,

a,b,c= cavity integers, only one which can be zero, and

i,w,h= length, width, height of the screenroom, m

The resonant characteristics of a shielded enclosure are different for

each test in which the equipment under test in the enclosure is physically

different or in a different location. Errors as great as 50 dB may be

4.2
present in screenroom field strength measurements , depending on frequency

and the characteristics of the individual enclosure.

Stuckey, Free and Robertson [4-1] have investigated the near-field

effects on measurement accuracies in a shielded enclosures . They have

postulated a theory "which accounts for the observed inaccuracies in terms

of a near- field measurement effect which is present in a shielded enclosure

environment and absent in the corresponding open-field environment". This near- field coupling hypothesis is used to explain the measurement

differences between the open-field and screenroom configurations at

frequencies below the first resonance of the enclosure. According to their

hypothesis, "the coupling between two antennas includes inductive coupling

and capacitive coupling as well as the normal coupling involving the

radiated field" . Their data show a deviation of about ± 40 dB between the

open-field and screenroom environment. Their evidence supports their near- field coupling hypothesis. However, they suggest that additional

investigation of the near- field coupling phenomena is needed.

Krstansky and Standley [4-3] developed a theory of EMC measurements

in screenrooms. They applied electromagnetic theory to the problem of

obtaining reliable and meaningful screenroom EMC measurements . They divided

the electromagnetic problem of determining the field distribution inside a rectangular enclosure, into three separate and unique parts depending on the

operating frequency and enclosure dimensions, as follows:

Case 1, u> < u>-^, where u>-y is the lowest resonant frequency

Case 2, w = w, , where u>, is a given resonant frequency

Case 3, w > w, , for k sufficiently large.

Their conclusion for case 1 was that "the size of the enclosure, as well as source and receiver locations, is extremely critical. Hence a specification as to the allowable field strength at a given distance from an equipment under test is virtually meaningless unless the screenroom

4.3
dimensions are included in the specification" . They consider case 2 to be

the most serious problem area in that as w approaches some w, value , the

fields become quite large. The exact value of u>. depends primarily on the

screenroom dimensions and location of objects within the enclosure, and

measurements made in one enclosure do not necessarily repeat in another

enclosure. For case 3 the density of w, values become so large that single

mode resonances are usually not observed and the theoretical field

distribution becomes very complex. "A posssible solution to this problem is

a statistical procedure involving mode mixing techniques". However, "a

detailed investigation of this approach is required to prove its superiority

over current methods".

In making EMC measurements, the antenna factor of any antenna varies

with the conditions under which the antenna is used [4-4] . Antenna factor

is defined as the ratio of the standard ( known) calibrating field to the rf

voltage delivered to the receiver by the antenna being calibrated when the

antenna is terminated in a resistance of 50 ohms. It is given by the

eq

K= f- , (4-2)

50

where E = the electric field strength incident upon the

receiving antenna, and

V -= measured voltage at the input of the 50 Q

measuring device to which the antenna is connected.

Usually the antenna factor is given in dB and written in the form

K,dB= 20 log(f—). (4-3)

50

Eqs (4-2) and (4-3) do not take into consideration the angles of

incidence and polarization diversity of the received signal with respect to

the transmitted signal. Mutual coupling between antennas should be accounted

4.4
for when the antennas are closely spaced. In a screenroom there are multiple

reflections due to the walls, floor, and ceiling. In addition, the mutual

impedance between the receiving and transmitting antennas may be

significant. Finally, resonances and nulls caused by enclosure interaction

with the antennas make the error of using the ideal free -space antenna

factor unreasonably large.

4 . 2 Measurement of Antenna Factors in a Screenroom

To determine antenna factors in a screenroom, the two antenna method as described in MIL-STD-461A was used. Figure 4-1 shows the instrumentation and

configuration used for making the measurements . This procedure requires two

antennas with "identical" characteristics (identical loading effects, and

equal impedance and gains). However, the observed antenna gain is a function

of position in the screenroom; hence the measured antenna factor is also a function of position.

In the two -antenna method, the transmitter output level is adjusted to

obtain a received signal, V , from the receiving antenna. The receiver and

K. signal generator cables are then disconnected from the antennas and directly

connected to each other to get the direct signal, V . The ratio of these

two levels is used to find the "realized gain" '^ or effective gain, '^ G ^-^ of eff

the two identical antennas being calibrated using the eq

G^^^=(47rd/A)(V^/V^), (4-4)

where d= distance between the two antennas shown in figure 4.1, m,

V = radiated signal received by the receiving antenna, V,

R

V = direct signal of the transmitted antenna via the

coupled cables, V,

A = 300/f,MHz, = free -space wavelength, m.

It should be noted that a measurement of "true gain" requires a knowledge of antenna impedance or that the antennas and loads be conjugately

4.5
matched in impedance, with lossless matching networks. However, after

measuring the "effective gain" by the two antenna method, the antenna factor

is calculated using the eq [4-5]

K= 9.73/(aJg^^^). (4-5)

Substituting for G from eq (4-4) into (4-5) one gets,

K= 9.73/2j7rAd(V^/V^)

Substituting (300/f,MHz) for X and taking 20 times the log of K, one gets,

(f, MHz)(V)

K,dB= 10 log[ ^^^^^ ^ —] -16. (4-6)

R

Using eq (4-6) and the instrumentation of figure 4-1, one can determine an

"in situ" antenna factor for two identical antennas at two specific

locations in a given shielded enclosure

.

4.3 Measurement of Antenna Factors at an Open Field Site

To determine antenna factors at an open field site, the two techniques

normally used are the "standard field" and the "standard antenna" method. At

NBS the standard antenna approach is used for calibrating dipoles or similar

antennas in which the horizontal component of electric field is to be

determined. The standard field method is used for calibrating monopoles or antennas in which the vertical component of electric field is desired.

4.3.1 Standard Antenna Method

The antenna factors are normally determined at an open field site. The

calibration is performed by radiating a field of convenient magnitude using

a transmitting half-wave dipole or a log periodic antenna (at frequencies

above 80 MHz) [4-6, 4-7]. The calibrations are done at a separation

distance of at least 2A between the transmitting and receiving antennas. The

receiving antennas are normally set at a height of 3 meters. The

4.6
transmitting antenna voltage is held at a constant level. The electric field

is detected by the standard A/2 dipole. The antenna being calibrated is then

substituted, in the same location, and its response recorded.

The NBS standard receiving dipoles used to measure the magnitude of

electric field strength consist of self -resonant dipoles with a high

impedance balanced voltmeter built across the center gap of the antenna. The

balanced voltmeter uses a Schottky diode detector, an R-C filter network,

and a high resistance transmission line. The dc output of the filter is

measured with a high impedance (100 MQ) voltmeter.

The antenna voltmeter dc-to-rf voltage transfer function is measured,

and from the dc output of the antenna the resulting rf voltage at the

essentially open-circuit dipole terminals can be determined. The calibrating

field strength at the measurement location is obtained from the eq

eff

where E = strength of the standard field, V/m,

V = open-circuit rf voltage of standard dipole, V, oc L _-= standard dipole effective length, m,

= (A/tt) [tan(7rL/2A)] , A = free -space wavelength, m, and

L = length of standard (self resonant) dipole, m.

Antenna factor is defined as the ratio of the standard calibrating

field to the rf voltage delivered to a 50 ohm receiver by the antenna being

calibrated. This antenna factor (K) , expressed in dB , is defined in

eq (4-3) and is determined from the standard field in eq (4-7). Measuring

4.7
the rf voltage and using eqs (4-7) and.(4-3) one can obtain the antenna

factor from

V

K, dB = 20 log(E /V ) = 20 log[

""^ ]. (4-8)

^ ^" ^50 eff

4.3.2 Standard Field Method

The antenna factors of vertical monopoles are calibrated by immersing

the receiving monopole antenna in a known field above the NBS groundscreen

and about 20 meters from the transmitting vertical monopole . A block

diagram of the instrumentation used for this type of calibration is given in

figure 4-2. The receiving antenna output at the end of the transmission line

is applied to a calibrated 50 ohm spectrum analyzer having a VSWR < 1.05.

This spectrum analyzer is used as a frequency- selective voltmeter to measure

the received rf voltage (V^.^^) at each frequency tested.

The calibration of vertical monopole antennas at NBS is done at an outdoor field site which has approximately a 30 x 60 meter conducting mesh

stretched over a concrete slab. The strength of the calibrating field is

calculated at each frequency in terms of the base current in a thin

transmitting vertical monopole antenna. The vertical component of E field,

as a function of base current and position, is given by the eq

301 -j/3r- -j/3r. -j^r

o e 1 e 2 e o E = [ . , ° J[- +- -2cos(l3h)(- -)] (4-9)

o sin(y5h) r- r^ ^A' / \ j, /

j

where E = vertical component of standard E field, RMS V/m,

I = monopole antenna base current. A,

h = height of transmitting monopole, m,

/3 = 27r /wave length in meters,

r- = distance from top of transmitting monopole to field

point, m, r_ = distance from image of top of monopole (in ground) to

field point, m.

4.8
r = distance from bottom of monopole to field point, m.

Eq (4-9) is number (10-72) on page 323 of Jordan [4-8].

The base current of the transmitting monopole is measured directly or obtained from the eq

^o = t

(^-1°)

m

where

I = monopole current, A,

o V = measured base voltage of the transmitting monopole, V,

Z. = magnitude of the monopole input impedance, f^.

At frequencies up to 30 MHz, a 2.5 m transmitting monopole is used to

generate a standard field. At frequencies above 30 MHz, the monopole height

is reduced to a value of A/4. At frequencies below one -tenth of the selfresonant A/4 frequency, the radiation resistance is negligible and the input

impedance is a capacitive reactance. The capacitance of a thin electricallyshort whip can be calculated from the eq

55-^63^L (4-11) ln(h/a)-l ' where C = monopole input capacitance, pF,

h = monopole height, m, a = monopole radius, m. The above eq is number (11) on page 306 of Schelkunoff [4-9].

At frequencies above one -tenth the self -resonant frequency, the input

impedance of the monopole (Z . ) is calculated from Schelkunoff ' s "mode"

theory using eq (108), page 433 of reference [4-9]. The impedance values of

this eq have been verified for the 2.5 meter whip by measuring with a commercial vector impedance meter. Eq (4-10) has been experimentally

verified at frequencies near self -resonance by measuring the transmitting

monopole base current with a calibrated vacuum thermocouple. Another check

of calculated E field has been made at the resonant frequency with a small

4.9
active probe which was calibrated in the known field of a TEM cell. The rf

voltage on the transmitting monopole ranges from 3 volts at self -resonant

frequencies to 300 volts at frequencies well below resonance.

The antenna factor (K) , in dB , is determined from the eq

K = E -V, (4-12)

o where E = field strength calculated from eq (8), dB/zV/meter, V

= pickup rf voltage delivered to the calibrated 50 ohm

spectrum analyzer, dB/zV.

4.4 Experimental Measurements of Antenna Factors

Two types of antennas were used for the measurement of antenna factors;

monopoles (104 cm rods) and broadband dipoles (biconicals) . These antennas

are used for emission and susceptibility tests at frequencies from 10 kHz to

200 MHz . The antenna factors of the monopoles and broadband dipoles were measured at an outdoor field site using the standard techniques developed by

NBS for calibrations. Additional measurements of the two types of antennas were made in the NBS screenroom using the MIL-STD-461A two antenna method.

All of the antenna factor values for the antennas are given in graphical

form, as described in the following section.

4.4.1 Biconical Dipoles

Figure 4-3 is the antenna factor graph given in the manufacturer's

manual for a typical biconical dipole. It is seen that the antenna factor as

a function of frequency varies from a minimum value of 7 dB, at its

resonance dip, to a maximum value of 18 dB . Figures 4-4 to 4-6 show antenna

factor curves as given in the antenna manufacturer's manual. Figure 4-4 is

for an antenna separation distance of 1 m, figure 4-5 is for a separation of

3 m, and figure 4-6 is for use in the far -field. All three figures are probably intended by the manufacturer for use at an open field site (free

space) because the antenna factor variation is only from 4 dB (minimum

value) to 18 dB maximum. Figures 4-7 and 4-8 give the far-field values of

antenna factor measured at NBS for the 2 biconical antennas purchased from

4.10
the manufacturer. These were calibrated at the outdoor NBS field site, using

the standard antenna method, with each antenna located at a distance of 15 m from a transmitting dipole. The height above ground was 3 m for both

transmitting and receiving antennas. A height of 3 m is theoretically

sufficient to reduce the ground proximity effects to a negligible value. As

shown in figures 4-7 and 4-8 , the two biconical antennas have measured

values of antenna factor which are nearly identical and can thus be used

legitimately as an identical pair for making two-antenna gain calibrations

as described in section 4.4. The total range of measured antenna factor is

from 4 dB (minimum) to 17 dB (maximum)

.

Figure 4-9 is a sketch of the screenroom layout for measuring the gain

(G _-) of two biconical antennas in the NBS screenroom. The two antennas eff were located on the center line (long dimension) of the screenroom with both

antennas at a height of 1 m above the floor. The dipoles were parallel to

each other and adjusted for horizintal polarization as indicated in the

figure. Two different separation distances were checked, 1 m and 3 m. Figure

4-10 gives the two antenna factor curves for the separation distances of 3 m and 1 m. The effect of screenroom resonances and reflections on the

measurement of antenna factor for a separation distance of 3 m is from -2.3

dB (minimum) to 54.9 dB (maximum). The range for aim separation distance

is 4.8 dB (minimum) to 50.1 dB (maximum). The reduced variation at a closer

separation distance is apparently due to reduction of screenroom (cavity) Q

by the close proximity of the receiving antenna with its associated power

absorption.

Figure 4-11 shows a sketch of the screenroom layout with the two

biconical antennas spaced 1 m apart but with their centerline rotated 90

degrees in the screenroom. Figure 4-12 shows another room configuration

which was tested for the effect of screenroom reflections on measured

antenna factor. Figure 4-13 gives the measurement results for the three

screenroom configurations of figures 4-9, 4-11, 4-12 The data of figure 4-13 were obtained with a separation distance of 1 m between the antennas. The

4.11
data clearly indicate the screenroom resonance effects and multipath

reflection variability as a function of antenna placement.

Figure 4-14 compares the difference in antenna factor in dB for a biconical antenna as measured in a screenroom and on the open field site.

This graph indicates that using the open- field- site antenna factors as permitted in MIL-STD-461A will give a greater error in field strength

measurements in a screenroom than using the antenna factors as measured insitu.

Using the NBS designed EFM-5 [4-10], calibrated in a TEM cell [4-11],

the electric field at the receiving antenna location was measured in

conjunction with measurement of the antenna factor of the biconical antenna

by the two-antenna method. Using this antenna factor, the field was determined from eq (4-12) and compared with the applied field as measured

with the EFM-5. This applied field was kept constant while making the

antenna factor measurements. Figure 4-15 is a plot of the difference between

field strength measured with the calibrated EFM-5 probe and that measured

with the biconical antenna calibrated by the MIL-STD-461 two-antenna method.

The plot indicates that the electric field as determined from the measured

screenroom antenna factors can differ as much as 7 dB from the electric

field measured with a calibrated EFM-5 probe, over the frequency range of 20

to 140 MHz.

4.4.2 Vertical Monopoles

Figure 4-16 is a typical graph of antenna factor for a vertical

monopole as given in the manufacturer's manual. Two of these antennas were calibrated at the outdoor NBS field site, using the standard field method.

Each antenna was located at a distance of 20 m from the transmitting

monopole. The height (length) of the transmitting monopole was 2.45 m. The

receiving monopole base was 1 m above ground. Figure 4-17 shows that the two

calibrated monopoles have measured antenna factors which are nearly

identical and can thus be used for determining antenna factor by the two

4.12
antenna method. The antenna factors measured on the ground screen varied

from about 12 dB (minimum) to about 42 db (maximum)

.

The gains of the two vertical monopoles were measured in the screenroom

using the two antenna technique as described previously for the biconical

antennas. Again, two separation distances of 1 and 3 m were used. Figure 4-

18 gives the two antenna factor curves for separation distances of 1 and 3

m. The antenna factor range for the 1 m separation is 10.3 dB (minimum) to

55.2 dB (maximum). The range for the 3 m separation is 24.9 dB (minimum) to

86.8 dB (maximum).

Figures 4-9, 4-11, and 4-12 show a sketch of the screenroom layout with

the two biconicals spaced 1 m apart to test the effect of screenroom

interaction on the antennas. Using figures 4-9, 4-11, and 4-12 for reference

placement, the two vertical monopoles were placed in the same position as those occupied by the biconicals. Figure 4-19 gives the measurement results

of the three screenroom configurations. The figure demonstrates the

screenroom nulls and coupling effects inside the enclosure as a function of

antenna placement.

4.5 Discussion of Antenna Factor Measurement Accuracy in Shielded Enclosures

Figures 4-10, 4-13, 4-18, and 4-19 show that measurements in a screenroom environment differ significantly from measurements made at an open field site (figures 4-3, 4-4, 4-5, 4-6, 4-7, 4-8, 4-16, and 4-17).

Variations of antenna factor for frequencies at or above the first resonance

of the enclosure can be attributed to the screenroom resonances and to

multi-path reflections in the screenroom. Variations of antenna factor at

frequencies below the first resonance of the screenroom, where the

dimensions of the enclosure are small compared to the wavelength, are caused

primarily by coupling of the two antennas via the enclosure walls and normal

coupling of the two antennas due to the radiated field.

4.13
4.6 Error Bounds of Screenroom Measurements

The data presented above show the variability of measured antenna

factors in a screenroom as a function of frequency, antenna placement, and

distaijice' between the antennas. The error can vary from to 40 dB as a function of any of the parameters given in the previous sentence. The

measurements also indicate that antenna factors determined in a screenroom

by the MIL-STD-461A two -antenna method are not an accurate parameter for

subsequently measuring an electric field. The errors in the measured field

strength inside a screenroom using the antenna factor eq of MIL-STD-461A can

be greater than ± 7 dB for biconical antennas in the 20 to 140 MHz frequency

range

.

4.7 Conclusions

Large measurement errors can occur in a screenroom because of room

resonances, multiple reflections, distorted radiated fields, and errors in

the antenna factors as measured by the MIL-STD-461A technique. It has been

shown that measurement results depend greatly on the size of the enclosure,

location of the test setup in the screenroom, and location of the antennas.

Field strength measurements made with the calibrated antennas in the

enclosure may have large errors. The correlation of these measurements, to

open-field or other shielded enclosure measurements with different

dimensions and geometries, is small.

Determination of an in- situ antenna factor in a screenroom or using

antenna factors which have been determined in a free space environment do

not lead to accurate field strength measurements in a screenroom. To

establish accurately-known electromagnetic fields in a screenroom, a transfer standard such as a calibrated EFM-5 probe should be used to check

and measure the field. These small transfer probes do not interfere with or distort the electic field being measured [4-10].

4.14
Further theoretical and experimental work needs to be done in the

measurement of radiated EM fields , such as loading the screenroom with

absorbing material in a systematic manner and studying the behavior of the

dampened waves

.

4.8 References

[4-1] Stuckey, C.W. Free, W.R. , Robinson, D.W. Preliminary interpretation of

near-field effects on measurements, lEEE-EMC Symposium Record, June

1969, Asbury Park NJ

.

[4-2] Free, W.R. Radiated EMI measurements in shielded enclosures, lEEE-EMC

Symposium Record, July 1967, Washington DC.

[4-3] Krstansky, J.J., Standley, R.D. Theory of RFI measurements in shielded

enclosures, 8th Annual IEEE Symposium on EMC, Jan. 1966, San

Francisco, Ca.

[4-4] Bennett, W.S. Antenna factors in EMC measurements, IEEE International

symposiunm on EMC, Arlington Va. , 1983

[4-5] MIL-STD-461A; Military Standard, Electromagnetic Interference

Characteristics, requirements for equipment; pp 10-11, August 1, 1985

[4-6] Greene, F.M. NBS field- strength standards and measurements (30 Hz to

1000 MHz). Proc. IEEE, Vol. 55, pp 970-981; June 1967.

[4-7] Taggart, H.E. and Workman, J.L. Calibration principles and procedures

for field strength meters (30 Hz to 1 GHz) . National Bureau of

Standards Tech Note 370; March 1969.

[4-8] Jordan E.G. Electromagnetic waves and radiating systems, PrenticeHall, Inc. , 1950

4.15
[4-9] Schelkunoff S.A. and Friis, H.T. Antennas, theory and practice, John

Wiley and Sons, Inc., 1952.

[4-10] Larsen, E.B. , and Ries F.X. Design and calibration of the NBS

Isotropic Electric-Field Monitor (EFM-5), 0.2 to 1000 MHz, U.S. Nat.

Bur. of Standards, NBS Tech. Note No. 1033, 104 pages March 1981.

[4-11] Crawford, M.L. Generation of standard EM; fields using TEM

transmission cells, IEEE Trans. Electromag. compat., EMC -16 No. 4, pp

189-195 November 1970

4.16
Signal

Gen.

50

Output

Transmitting

Antenna

,

Receiving

Antenna

V

d= 1 m

or 3 m

10 dB

PAD

10 dB

PAD

I ^ >

1 Cable Connector

Screenroom

RCVR

50

Input

Figure 4-1. Test setup required by MIL-STD-461A for determination of

antenna factors

.

Mj

D.C. Voltmeter

Transmitting

Monopole

Rod Antenna

to be Calibrated

Ground Screen IZ> Calibrated

50 A Receiver

^rmocouple JMMConcreie Slab WMMmmMMm

I

R.F. SourceR.F. Voltmeter

.Added Capacitance

at Low Frequencies

Underground Transmitting Room

Figure 4-2. Instrumentation used at NBS open-field site to calibrate

monopole antenna factors

.

4.17
80 100 120 140 160 180 200

Frequency, MHz

Figure 4-3. Antenna factor of a typical biconical dipole taken from

the manufacturer's manual for a far- field separation

distance

COS 15-

<

100 120 140

Frequency, MHz

200

Figure 4-4. Antenna factor of a typical biconical dipole taken from the

manufacturer's manual for a separation distance of 1 m.

4.18
20 40 60 80 100 120

Frequency, MHz

140 160 200

Figure 4-5. Antenna factor of a typical biconical dipole taken from the

manufacturer's manual for a separation distance of 3 m.

80 100 120 140

Frequency, MHz

180 200

Figure 4-6. Antenna factor of a typical biconical dipole taken from

the manufacturer's manual for a separation distance of

15 m and an antenna height of 3 m.

4.19
20 40 60 80 100 120 140

Frequency, MHz

160 180 200

Figure 4-7. Antenna factor of Biconical No. 1 for a "far field"

separation distance, as measured at NBS.

20 40 60 80 100 120 140 160 180 200

Frequency, MHz

Figure 4-8. Antenna factor of Biconical No. 2 for a "far field"

separation distance, as measured at NBS.

4.20
Transmitting

Biconical

Antenna

d = 1 m Note: height of antenna

(or 3 m) above floor = 1 m.

Receiving

Biconical

Antenna

1.7 mCoax Cable

Figure 4-9. Placement of biconical antennas in NBS screenroom,

position 1.

4.21
60

1 Meter spacing

3 Meter spacing

-10 1

20 40 60 80 100 120 140 160 180 200

Frequency, MHz

Figure 4-10. Antenna factors for blconical dipole calibrated in the

NBS screenroom at spacings of 1 and 3 m.

4.22
Copper Clad Table

Coax Cable

Figure 4-11. Placement of biconical antennas in NBS screenroom,

position 2.

4.23
Figure 4-12. Placement of biconical antennas in NBS screenroom,

position 3.

4.24
80 100 120 140 160 180 200

Frequency , MHz

Figure 4-13. Measurement results of three screenroom placements with

an antenna separation distance of 1 m.

4.25
14

12

m

T3 10

a>u

c 8

0)

1^9i 6

O

k. 4

^•» O

2

COc

1-2

-4h

-6

100 120 140 160 180 200

Frequency, MHz

Figure 1-11 Difference between antenna factors measured in a screenroom and on the NBS field site for a biconical

dipole.

60 80 100

Frequency, MHz

Figure 1-15. Difference between field strength measured with a calibrated transfer probe (EFM-5) and that measured

with a biconical antenna calibrated by the MIL-STD-161A

two-antenna method.

4.26
70

m 60 ^ 50

(T3

a>

1

CtO a <D LU CU}

XJ u LU

Lim "O

CD c

m

a;c

< 30

20

DO

LU

o

c

15 20

Frequency, MHz

25 30 35

Figure 4-16. Antenna factor for typical 1.04 m vertical monopole

taken from the manufacturer's manual.

50

45

40 ff10

Free Space Antenna Factor of Vertical Monopoles

Monopole 1

Monopole 2 «>

10 20

Frequency, MHz

30 40

Figure 4-17. Free-space antenna factor of two vertical monopoles as calibrated at the NBS open-field site.

4.27
15 20

Frequency, MHz

35

Figure A- 18. Antenna factor for 1.04 m vertical monopole calibrated

in the NBS screenroom at different antenna spacings.

90

25 30 35

Figure 4-19

5 10 15 20

Frequency, MHz

Antenna factor for 1.04 m vertical monopole calibrated

in NBS screenroom at different antenna locations for a

1 m antenna separation.

4.28
5. Measurement of Broadband Impulsive Signals

5.1 Introduction

The impulse bandwidth (IBW) of an EMI receiver is used for measurement of

broadband interference. The IBW may be determined by several different

approaches. The spectrum amplitude of an impulse generator (IG) as a function

of frequency, S{f), may also be measured by several approaches. Four of these

possible methods were evaluated at NBS for comparison purposes. The four

techniques are defined and discussed in this chapter.

An RF signal is considered to be narrowband or broadband depending on whether its frequency spectrum is less than or greater than the bandwidth of

the receiver. The overall bandwidth of a receiver is determined essentially

by the selectivity of the IF amplifier strip and is typically of the order of

0.1 percent of the receiver tuned frequency. The term "cw" stands for a single-frequency sinusoid of constant amplitude. The term "impulse" is used

for a baseband (DC) pulse or an RF burst (pulsed RF) whose time duration is

short compared with the period of the receiver circuit which responds to the

impulse. This period is approximately the reciprocal of the IF bandwidth for

a well -designed EMI receiver.

A common practice is to divide broadband interference into two categories, which may be defined as follows: (1) Random noise consists of a large

number of overlapping signals with random amplitude, frequency, and relative

phases. Examples are thermal noise and shot noise. (2) Impulse noise

consists of a single short pulse of voltage, or repetitive pulses, causing a signal with systematic relative phase and amplitude vs frequency. Examples

are switching transients, automobile ignition noise, and interference from

radar transmitters. The response of a receiver to either type of broadband

signal is determined, partially, by the receiver bandwidth. This is because

all frequency components of the input signal which fall outside the receiver

bandwidth are rejected, or at least amplified less than the frequency

components at the center of the passband.

Differences between the possible types of RF signals can be illustrated

by comparing receiver response (indicated level) as a function of receiver

bandwidth. For a cw signal, the response amplitudes for a peak-voltage,

average-voltage, or RMS-voltage detector are independent of the receiver

bandwidth. For random noise, the average voltage and RMS response values of a

5.1
receiver are proportional to the square root of bandwidth. For impulse noise

of low repetition rate, the average voltage is typically independent of

bandwidth but the peak value is proportional to bandwidth (among other

things). This proportionality factor makes it convenient to define an impulse

bandwidth for a receiver. Actually, the response of a tuned receiver to an impulse, as measured with a peak-reading detector, depends not only on bandwidth but also on the shape of the bandpass curve and phase

characteristics of the receiver. The response of ideal interference meters to

various inputs, as a function of bandwidth, is summarized in table 5-1.

In most common approaches to impulse measurement, the impulse bandwidth

of the receiver is a calibration factor determined by adjusting the output of

a cw generator to give the same indication on the receiver output meter as the

peak response to an impulse generator of known characteristics. The impulse

bandwidth in MHz is then the output of the cw generator in yV, divided by the

output of the impulse generator in yV/MHz. Some commonly used definitions

relating to broadband interference are discussed in the next section.

5.2 Calibration of an EMI Receiver and Measurement of an Impulse

Generator Using the Video Pulse Technique of MIL-STD-462

The video pulse technique involves application of a train of (nominally)

identical repetitive pulses to an EMI receiver which has been designed with

the "proper" shape of passband. This means that the receiver acts as a filter

having a well-rounded (Gaussian shaped) passband, rather than a rectangular

passband. The measurement approach is prescribed by MIL-STD-461/462 [5-1,5-2]

and is unique in the sense that the EMI receiver itself is used to measure the

impulse bandwidth of the receiver. That is, no calibrated external pulse

standard is required to calibrate the receiver IBW. As specified in the MILSTD, the area of the receiver response envelope and maximum envelope amplitude

are each measured. From these measurements the receiver IBW and spectrum

amplitude S(f) of the IG are determined. The calibrated receiver (with a calibrated antenna) can then be used to measure radiated impulsive interference. A sketch of the intermediate frequency (IF) output voltage v(t) of a typical EMI receiver is given as follows.

5.2
v(t)

ENVELOPE = V(t) V (t)

EQUAL AREA

- t In the above sketch:

v(t) = receiver IF voltage output in the time domain, RMS volts,

V(t) = envelope of the receiver voltage response curve, RMS volts, and

%{t) = maximum voltage value of the envelope of the receiver response

curve, RMS volts.

Definitions of the terms involved are given as follows:

a. Receiver impulse bandwidth

v.(t)

(IBW, Hz) =

or (IBW,Mhz) =

_ Maximum envelope height, volts

r

"^ w/.\.. ~ Envelope area, volt»sec

* Maximum envelope height, cm (Envelope area, cm^) (Scope sweep speed, sec/cm)

Maximum envelope height, cm (Envelope area, cm^) (Scope sweep speed, ysec/cm)*

b. Impulse spectrum amplitude

CO <iif\ \l^^^r.lu-. - /-i/r \ ( \ii^r\A^ En velope area, volt'sec S(f), Volts/Hz = (1/Go) i V(t)dt = — RCVR gain

*

Also

'^{i\ Vnn-c/H7 - m^ ' _ Maximum envelope height, volts

:>^t;, voils/hz - (Q^)(igw) - (rcVR gain, Go)(RCVR IBW, Hz)' or

<^/fx \ii\Aw _ (Maximum envelope height, cm)(Scope gain, yv/cm) ^^' ^^^"^ (RCVR gain, Go)(RCVR IBW, MHz)

(5-la)

(5-lb)

(5-lc)

(5-2a)

(5-2b)

(5-2c)

_ (Envelope area, cm^)(Scope sweep speed, Ms/cm)(Scope gain, uv/cm) (^ oa\

RCVR gain, Gq

5.3
A block diagram of the instrumentation used at NBS for the MIL-STD-462

approach for measuring impulse bandwidth of a receiver and spectrum amplitude

of an impulse generator is given in figure 5-1. The EMI receiver used for our tests has a three-position bandwidth switch, with selectable bandwidths of 10

kHz, 100 kHz, or 1 MHz. This bandwidth switch was set in the 1 MHz position

for all the data reported here. At each receiver tuned frequency tested, the

receiver "CAL" knob (gain control) was adjusted to obtain a receiver dial

indication of 40 dByV, with the receiver step attenuator in the 40 dB position. This corresponds to a voltage level of 80 dByV = 10 mV, which is the

true receiver input signal level as determined by the calibrated voltmeter

(1 V minus 40 dB = 80 dBpV). The overall cw gain of the receiver, Gq, was determined by measuring the IF output with the oscilloscope shown in figure

5-1, and dividing this voltage by the 10 mV RMS input to the receiver. That

is,

r - (cw peak-to-peak volts at IF output)/f 2/7] ,^ -v ^0

- 10 mV • ^^""^^

As shown in figure 5-1, a commercially available impulse generator was used for our tests. It is specified by the manufacturer to produce a spectrum

output that is flat within ±1 dB from 60 kHz to 1000 MHz. The output

amplitude is switch selectable from 10 to 101 dByV/MHz (RMS) in one-quarter dB

steps. The IG was set to its maximum output level of 101 dByV/MHz for all the

data included in this report. The receiver input level for impulsive signals

should thus be 95 dByV/MHz, after the 6 dB insertion loss of the 50 o. power

splitter. Photographs of the receiver response envelopes (video pulses) are shown in figure 5-2. The data for the MIL-STD calibration of the EMI receiver

and impulse generator are given in tables 5-2 and 5-3. A set of data was taken at frequencies near the top and bottom end of each of the eight receiver

bands, plus measurements at the middle of the highest frequency band. It can

be seen that the MIL-STD measurements of S(f) are within one-half dB of the IG

nominal output value of 101 dByV/MHz at all the frequencies checked below 800

MHz. At higher frequencies the measured values of S(f) range from 101.9 to

103.6 dBpV/MHz.

The definition of impulse bandwidth of a narrowband filter (receiver) is

given in the IEEE dictionary of electronic terms [5-3] as follows: "The peak

value of the response envelope divided by the spectrum amplitude of an applied

5.4
impulse." It is seen that this definition agrees with the MIL-STD approach

[5-1,5-2,5-4], A copy of Section 4.2.3.5 and 4.2.3.6 of MIL-STD-461 is quoted

here for reference purposes.

Calibration of Measuring Equipment . A1 1 measuring equipment shall be

calibrated in accordance with MIL-STD-461.

Impulse Generators . Impulse generators shall be calibrated as follows:

(a) Apply the output of an impulse generator to the input of an amplitude-linear receiver having synchronously tuned, less than critically

coupled circuits. EMI receivers are satisfactory for this purpose if

their impulse bandwidth is at least five times the repetition rate of the

impulse generator. Any AGC system shall be disabled and the AGC line

firmly referred to ground with a low impedance voltage source of appropriate value.

(b) Obtain an oscilloscope pattern of the overall receiver response at

the IF output. The oscilloscope controls shall be so adjusted that the

pattern is as large as possible within the calibrated area on the face

plate. Either photograph or trace the pattern. Record the oscilloscope

sweep speed setting. (The sweep speed shall be calibrated accurately.)

(c) Use a planimeter or other integrating device to determine the area

of the pattern. This operation shall be carried out at least 10 times

and the average of the readings taken as the area.

(d) Calculate the impulse bandwidth of the receiver in accordance with

the following formula:

(Pattern height in cm)

IBW in Hz = (Pattern area in cm2) (Sweep speed in sec/cm)

5.5
(e) Connect a sinewave generator to the receiver and at f^ adjust the

output until the peak pattern height is the same as that obtained with

the impulse generator.

(f) Connect an rf power meter to the signal generator and measure the

power output. Calculate the rms voltage of the sinewave signal.

(g) The output of the impulse generator is now calculated by taking the

rms voltage calculated in step (f) divided by the impulse bandwidth

determined in step (d).

5.3 Measurement of an IG by the NBS FFT Calibration System

An impulse generator (IG) can be calibrated by measuring the train of

output pulses with a sampling oscilloscope and then taking the Fast Fourier

Transform (FFT) of the average time domain pulse. This approach is used as the basis for the NBS method of calibrating the spectrum amplitude of broadband impulse sources [5-5]. The technical literature relating to the term

"spectrum amplitude" is confusing due to the use of several different names

for the same quantity. Other names include spectral intensity, spectral

density, voltage spectrum, and impulse strength. The name preferred by IEEE

is spectrum amplitude, defined as double the magnitude of the Fourier

transform of the time domain pulse. That is, the spectrum amplitude vs frequency, S(f), is computed from the measured waveform of the time domain

pulse, v(t), using the basic IEEE definition [5-6], as follows:

S(f) = 2 |V(f)| (5-4)

where

V(f) = / v(t) e'^^^'^^dt. (5-5)

—00

The unit of S(f) is volt«second, or its equivalent volts/hertz. The

units commonly used by the EMC community are jjV/MHz or dBpV/MHz. The decibel

unit is defined as S(f) in decibels above 1 uV/MHz

20 log r S(f) in tiV/MHz

10 L 1 yV/MHz J5.6
The NBS and IEEE definition given by eqs (5-4) and (5-5) differ by 3 dB from

that used by the EMC community. Therefore, as will be seen in the measurement

data, a 3 dB correction factor was applied to the NBS FFT data in order to

compare these results with the MIL-STD-461 measurements. In the time domain,

voltage pulses are measured with respect to zero volts. In the frequency

domain, however, they are usually expressed in RMS voltages. For a sine wave,

the peak value is /? times the RMS value. This /7 factor causes a 3 dB

difference between the two voltages. Consequently, when a peak responding

receiver that has been calibrated in terms of the RMS sine-wave voltage is

used to measure S(f), the results will be 3 dB lower than if it were calibrated in terms of the peak value.

A block diagram of the NBS Automatic Pulse Measurement System (ARMS) is

given in figure 5-3. As shown, it consists basically of a sampling oscilloscope interfaced with a computer. The IG used in our tests does not provide

trigger output pulses for the oscilloscope. Therefore, a trigger pick-off

delay line was used, as shown in the figure. The 6 dB power splitter, 30 dB

pad, and 30 dB amplifier insure that the IG and delay line are matched to the

50 Q measurement system. The NBS ARMS is described in detail in [5-7] and

[5-8]. Briefly, NBS calibrates impulse generators by measuring the time

domain waveform of an average pulse from a train of repetitive pulses, and

then computes the frequency domain spectrum amplitude. The ARMS consists of

an 18 GHz bandwidth, 50 o. sampling oscilloscope interfaced with a computer.

The time domain waveform is digitized and stored in the computer memory. By

acquiring many waveforms and signal averaging, it is possible to improve the

signal-to-noise ratio of the measurement. The spectrum amplitude is then

computed, using the fast Fourier transform, eqs (5-4) and (5-5).

Figure 5-4 is a photograph of the oscilloscope trace of the IG output

pulse and a plot of the corrected pulse as supplied to the computer for FFT

analysis of the spectrum amplitude. Table 5-4 is a list of the computed

spectrum amplitude values as measured in the frequency range of 50 to

2400 MHz. Figure 5-5 is a graph of the data for frequencies up to 2 GHz. It can be seen that the measured values agree with the specified IG output of

101 dByV/MHz within about 1 dB for all frequencies up to 1 GHz.

5.7
5.4 Approximations of IG Output in Terms of the Measured Area of the Pulses

Certain "idealized" types of pulses can be described mathematically and

their spectrum amplitude can be readily calculated. The signal output of

actual impulse generators approximates these ideal waveforms quite closely.

This approach is used by many manufacturers of IGs. An example is the IG

purchased for this project. Such "standard" IGs can be used to calibrate the

IBW of a receiver, and are similar to the IG built into the usual field

intensity meter for self calibration. That is, the receiver gain control is

adjusted to obtain the correct indication (in dByV/MHz) at each receiver-tuned

frequency.

A common type of receiver IG uses a pulse forming transmission line of

known electrical length that is charged to a known (regulated) DC voltage.

The repetitive transient spikes produced by discharging the line have a time

duration of about one-half nanosecond each, and a repetition rate of typically

60 Hz. The spectrum of an IG having short (rectangular) pulses is quite flat

at frequencies up to about (0.2/Tq), with an amplitude given approximately by

[5-9]

S(f) = 2A = 2 Vq. To (5-6)

where: S(f) = spectrum amplitude of the IG, volts/hertz,

A = area of each pulse, volts»seconds,

Vq = DC charging voltage on the line, volts, and

Tq = pulse duration, seconds.

The above approximation, that S(f) is equal to double the area under the

time domain curve of baseband pulses, is strictly true only for idealized

delta-function impulses in which the pulse duration is zero and the pulse

height is infinite. In this case the frequency spectrum V(f) has constant

amplitude, independent of frequency. For pulses of finite duration, S(f) at

low frequencies is also approximately twice the pulse area. Actually, the

equation for the frequency spectrum of rectangular pulses is

s(n = 2 vjo

sin(TT f Tq)

(5-7)

The equation for the frequency spectrum of triangular pulses is

5.8
S(f) = 2 VqTo

sin(iT f Tq)

(tt f To)

(5-8)

Figure 5-6 shows graphs of both the time domain and frequency domain

functions for three common (ideal) types of pulses. For short pulses and low

frequencies, the (sin x)/x function approaches 1, and S(f) is approximately

2 Vq Tg as given by eq (5-6), The IBW of a receiver to be used for measuring

impulsive interference can then be calibrated with a "standard" IG in terms of

the basic equation

V (t) m^ ' V (t)

m^ ' ^^W = G^STfy =

2 GoVJo ' (5-9)

where G^ = receiver system tuned-frequency cw gain.

The theoretical, approximate spectrum amplitude of the commercial IG, at

frequencies up to 1 GHz, can thus be calculated by several approximations, as follows:

a. Assuming that the IG pulses are rectangular, with a peak amplitude of V^

and a duration given by the time interval at the half-voltage point, but

omitting the overshoot and oscillatory part at the end of the pulse (see

figure 5-4), then

S(f) = 2A = (2)(0.23 V)(450 ps) + 50 dB attenuator pad

+ 6 dS power splitter - 3dB for the EMC definition

= (2)(0.23pV)(450 ys) + 53 d6 = 99.3 dByV/Mhz.

The above value can be compared with the approximate 100.5 dBpV/MHz value

measured with the NBS FFT system, as shown in figure 5-5, for frequencies

up to 1 GHz.

b. Assuming that the IG pulses are triangular in shape with a time duration

measured at the base of the triangle, but omitting the negative and

positive pieces of the pulse after the first triangle, then

S(f) = 2A = (0.23 V)(800 ps) + 53 dB = 98.3 dBpV/MHz.

c. Using the measured area of the first positive part of the pulse.

5.9
S(f) = 2A = 98.1 dByV/MHz.

d. Using the same approach as (c) above, but assuming that both the positive

and negative parts of the pulse should be added,

S{f) = 2A = 100.7 dByV/MHz.

The above value is in best agreement with S{f) as measured with the NBS

FFT system and with the nominal 101 value of the commercial IG tested.

Measurements of the built-in step attenuator of the IG were made for all

step positions, at a frequency of 300 MHz. The attenuator was found to

be accurate within ±0.2 dB.

5.5 References

[5-1] MIL-STD-461A. Military Standard, Electromagnetic Interference

Characteristics, Requirements for Equipment; 1968 August 1. Also Notice

1, 1969 February 7; Notice 2, 1969 March 20; Notice 3, 1970 May 1; Notice 4, 1971 February 9; and MIL-STD-461B, Notice 1, draft, 1983

January 5.

[5-2] MIL-STD-462. Military Standard, Electromagnetic Interference

Characteristics, Measurement of; 1967 July 31. Also Notice 1; 1968

August 1, Notice 2; 1970 May 1, Notice 3; 1971 February 9, and MIL-STD462A, draft; 1970 June 29.

[5-3] IEEE Std 100-1972. IEEE Standard Dictionary of Electrical and

Electronic Terms, the Institute of Electrical and Electronics Engineers,

Inc. New York: John Wiley & Sons, Inc.; 1972.

[5-4] MIL-STD-463. Military Standard, Definitions and Systems of Units,

Electromagnetic Interference Technology; 1966 June 9. Also, Notice 1;

1972 March 2.

[5-5] IEEE Std 376-1975. IEEE Standard for the Measurement of Impulse

Strength and Impulse Bandwidth. The Institute of Electrical and

Electronics Engineers, Inc.; 1975.

[5-6] Andrews, J.R.; Arthur, M.6. Spectrum amp! itude--definition, generation

and measurement. Nat. Bur. Stand. (U.S.) Tech. Note 699; 1977 October.

96 p.

[5-7] Gans, W. L. ; Andrews, J. R. Time domain automatic network analyzer for

measurement of rf and microwave components. Nat. Bur. Stand. (U.S.)

Tech. Note 672; 1975 September.

5.10
[5-8] Andrews, J. R. Impulse generator spectrum amplitude measurement

techniques. IEEE Trans. Inst. & Meas., Vol. IM-25, No. 4: 380-384; 1976

December.

[5-9] Larsen, E.B. Calibration of radio receivers to measure broadband

interference. Nat. Bur. Stand. (U.S.) NBSIR 73-335; 1973 September.

75 p.

Table 5-1. Response of an ideal receiver

to various types of input signals.

Type of

Detector Type of RF Signal

Sine Wave Random Noise

Periodic

Pulses

Average Receiver Independent of

(voltage) indication is: bandwidth (BW)

Peak Receiver Independent of

(voltage) indication is: bandwidth (BW)

RMS Receiver Independent of

(voltage) indication is: bandwidth (BW)

Power Receiver Independent of

indication is: bandwidth (BW)

Proportional Independent

to /BW of BW

Proportional

to BW

Proportional Proportional

to /BW to /BW

Proportional Proportional

to BW to BW

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C- <u OJ (U Ol ei> ^^ (U o o 0) E ea. ca. C3l <u •1— eu cn > > > > 1/1 fO > > 4J o o o •-» i- 4-> t-> cn o •r- •^ • r* •p- ^— c •f— t— r— •p* >< .C ^— r* r^ ^— 3 ^— p— m

+j 0) <u <D (U 3 (U •p« • f— eu fO cn eu (U <u 3 to 3 3 I- o o o O o ea. 'i u o u o E • r— > > > to 13 to to 0) j= cu 0) CU O) E o eu 1/) to <D eu c c c (U <U (U <u >

a. Qi Qi OH q; 1—1 z Q£ o o Qi 3 LU LU LU q: S q: Oi cC

5.13
Table 5-4. Spectrum amplitude of the commercial impulse generator,

measured with the NBS FFT calibration system.

S(f), dBijV/MHz

Frequency

MHz

s(f). dByV/MHz

Frequency

MHz

Using

IEEE

Definition

102.9

Using

EMC

Definition

99.9

Using

IEEE

Definition

102.6

Using

EMC

Definition

50 1250 99.6

100 103.5 100.5 1300 102.5 99.5

150 103.2 100.2 1350 101.5 98.5

200 103.5 100.5 1400 100.6 97.6

250 103.5 100.5 1450 99.2 96.2

300 103.7 100.7 1500 98.0 95.0

350 103.2 100.2 1550 95.7 92.7

400 103.5 100.5 1600 93.7 90.7

450 103.3 100.3 1650 92.0 89.0

500 103.4 100.4 1700 90.4 87.4

550 103.2 100.2 1750 87.7 84.7

600 103.6 100.6 1800 86.1 83.1

650 103.4 100.4 1850 83.1 80.1

700 103.7 100.7 1900 80.0 77.0

750 103.4 100.4 1950 81.7 78.7

800 103.4 100.4 2000 87.4 84.4

850 103.3 100.3 2050 87.0 84.0

900 103.9 100.9 2100 89.9 86.9

950 103.9 100.9 2150 89.1 86.1

1000 . 104.1 101.1 2200 89.0 86.0

1050 104.0 101.0 2250 82.3 79.3

1100 104.3 101.3 2300 78.1 75.1

1150 103.5 100.5 2350 79.6 76.6

1200 103.2 100.2 2400 83.1 80.1

5.14
SINE WAVE

(CW) SIGNAL

GENERATOR

COMMERCIAL

IMPULSE GENERATOR

(60 kHz - 1 GHz)

(Set to 101

yATTENUATOR

PAD (40 dB)

1

RF VOLTMETER^

ADJUST SIGNAL

FOR 1 VOLT

dbyV/MHz)

>r

POWER SPLITTER

TEE

(-6 dB) (-6 dB)

Calibrated Nominal

CW signal T, ^ J^ 95~dM/MHz

ANT. INPUT

COMMERCIAL

EMI RECEIVER

(1 MHz BANDWIDTH)

V

IF

ATTENUATOR

PAD (20 dB)

OUTPUT

>^

OSCILLOSCOPE .<

V

SYNC. PULSES

CAMERA

Figure 5-1. Instrumentation for testing the MIL-STD-461/462 approach for

measuring impulse bandwidth of a receiver and spectrum amplitude

of an impulse generator.

5.15
Figure 5-2. Photographs of the receiver impulse response envelopes (video

pulses) measured at NBS.

5.16
1

COMMERCIAL

IMPULSE

GENERATOR

ATTENUATOR

PAD 50 dB

(Set to 101

dbuV/MHz)

1-

POWER SPLITTER

TEE

ATTENUATOR

PAD (30 dB)

50 Q COAXIAL

DELAY LINE

AMPLIFIER

+ 30 dB

Trigger

V

SAMPLING

OSCILLOSCOPE

X in

:?'>:

D/A

^

t

Y out

A/D

±±.

COMPUTER

SAMPLING

HEAD (50 Q)

VDT AND

PRINTER

Figure 5-3. Instrumentation for the NBS automatic FFT pulse measurement

system.

5.17
a^mmm^-^l^m^^mm "m""^*' ^\^^««^^#a

Figure 5-4. Photograph of sampling oscilloscope trace of the IG output pulse

(top) and plot of the pulse supplied to the computer for the FFT

(bottom)

5.18
CMO

1 J^

1— V UJ

^^ ^- ^ i

P CO ^ i 1 l_

[ ^ U- \

u. ^ y to

. _i_. X CQ

H \

H

H

S

s

in

s'9

H

-s

o

c

f—<—I—I K—

H

O

SI

4-> E O 0)

4-> W> >,^

0)

1-

TO

O) E

+-> m

•— 3

Q. Q.

E

u. E uI. I/) U 2

O. 0)

«/> JC

4-> CJ

•—> -oc

«4-O

CNJ

I QV) I—

SI 00 a. I na _J

J- >-•

8

m

I

o»

ZHN/A^aP '(J>S

5.19
o o o o CVJ ^ vo 00 1 1 1 1

CQ o o o 13 CVJ ^ V£> 1 1 1 o

00I E

C-o

«•->cocu

nEO

"O

0)E o

h- O)

CVJ c=5 M

•r- ^—

Q. o E - «oE3

L. ^ o O

0)

II Q.

U)

OJ

tooco

CVI

0)(0 J*"""^ «>—

K-'l o o d

h- 1—

a ^- 14- M »= *= c ^ D

0> 1/1

•0 o1—o o >

<u CVJ oi o o

(— f—

a<to ^- >4- 3 )= ^ O.

i~ c ID

(/) 3

a> o c t—

•e o>c\J

0)

01 rHda

a)M

CO

0)c

to o

CVJI

CVJ

1= CVJ

O

I—o>CVJ

a; JZo

C/1I LD

O)

(-3

5.20
6. Overall Summary, Conclusions, and Recommendations

6.1 Field of a Parallel-Plate Transmission Line

MIL-STD-462 specifies the use of a parallel-plate transmission line

(stripline) having a plate spacing of 0.457 m (1.5 ft), an upper plate width

of 0.610 m (2 ft), and a lower plate width of 0.914 m (3 ft). This stripline

is used for making radiated susceptibility tests of electronic equipment in a known EM field. The characteristic impedance (Z^) depends mainly on the plate

spacing and width. It is also affected by its location, especially if the

line is inside a shielded enclosure (high-Q cavity). Therefore, the optimum

value of the terminating resistor for a stripline should be determined experimentally from tests of the E-field. The main criterion for a good standardfield setup is the level and uniformity of the generated field as a function

of frequency. MIL-STD-462 states that the expected Z^ value for this line is

83 n. Several equations are given in the literature for calculating 1^ of

parallel-plate lines. The line fabricated at NBS according to MIL-STD-462 has

measured values ranging from 80 to 160 n.

Measurements were also made of the input impedance of the stripline, with

the line terminated in two different values of resistance (83 ^ and 120 fi)

,

for frequencies ranging from 1 to 100 MHz. The data show that the input impedance varies with frequency for both values of terminating resistance. The

optimum value of terminating resistance appears to be in the neighborhood of

120 fi rather than the 83 n specified in the MIL-STD.

Measurements of electric field strength vs signal frequency were made for

two different locations of the stripline in the empty screenroom. The measured values of E-field at the center of the stripline, for both values of

terminating resistance, were within 2 dB of the theoretically calculated

values at all frequencies up to 40 MHz. The measured E-field at 30 MHz as a function of position along the line was correct within 2.3 dB and uniform

within ±0.5 dB over the 2 m length of the line. The effect of screenroom res- onances on the behavior of the MIL-STD stripline is clearly evident from the

measured excursions in impedance and field strength at frequencies above 40

MHz. These resonance effects would occur at lower frequencies if a larger

screenroom were used.

A parallel -plate stripline can be operated without a resistive termination at frequencies up to 30 or 40 MHz. When the line is open circuited, it

6.1
produces a field with high wave impedance (high E/H ratio). High level Efields can then be produced by connecting a broadband step-up transformer at

the input of the line. Also, a high level E-field can be produced with a low

power source if the line is resonated by means of an inductor. At higher frequencies (above 30 MHz) approaching self resonance of the system, it is necessary to measure the field strength at the calibrating point with a small

transfer probe that has been calibrated previously in a small stripline or TEM

cell. When a stripline or TEM cell is short circuited, it acts as a single

turn loop, producing a high level H-field with low wave impedance (low E/H

ratio).

6.2 Use of an Electrically Small Transfer Probe

An electrically small antenna (rf probe) can be used to test field level

vs frequency and position in a MIL-STD-462 stripline. Such transfer probes

can also be used to improve the construction and operation of other standardfield chambers by optimizing experimentally the value of terminating load

impedance. Similarly, the E- or H-field in a screenroom that has been con- verted into a long-wire chamber can be determined more accurately with a transfer probe than by relying on theoretical equations. Measurements for

this report were made with a calibrated NBS E-field probe (EFM-5) having

dipole antennas 5 cm in length. Because of its design and small size, it does

not perturb the field being measured.

6.3 Field of a Single-Wire Transmission Line

The long-wire antenna at NBS was installed in a shielded room. A No. 10

wire was suspended from the two end walls of the room, driven at one end and

terminated at the other. The dimensions of the NBS screenroom are 7 m long

X 3.7 m wide x 2.4 m high. The MIL-STD-462 equation for the characteristic

impedance is stated as usable when the distance of the wire to the ceiling is

less than one-third the room height. The calculated value according to this

equation is 422 a. The final experimental determination of the optimum value

was 380 n.

The field uniformity along the center line of the screenroom chamber was measured at 1 MHz, 15 MHz, and 30 MHz. The deviation with respect to the

field at the midpoint of the chamber was less than +0.3 dB at 1 MHz, less than

6.2
+1 dB at 15 MHz, and varied from +0 to -3 dB at 30 MHz. The data indicate

that an acceptable uniformity can be achieved in the long-wire antenna

chamber. However, terminating the antenna with the optimum resistance, as described in the MIL-STD, is necessary. The equation given in MIL-STD-462 was used to calculate the E field and compared with the measured values. The

vertical component was measured at several distances below the long-wire for

signal frequencies of 1, 2, 3, 5, 7.5, 15, 22, and 30 MHz. The distance below

the center of the wire ranged from 0.2 to 0.8 m. The measured field varied

both as a function of distance and frequency. The dB error in the calculated

field was greater as the distance from the center conductor was increased.

The data indicate that the average field strength produced in the long-wire

chamber is uniform within 3 dB at frequencies up to 30 MHz, provided the longwire is terminated in its characteristic impedance. Hoewever, it is recom- mended that a transfer probe be used to measure the field uniformity and

absolute value at any desired location.

6.4 Calibration of Antenna Factors for EMI Antennas

Measurements have been made at NBS of antenna factors in a screenroom,

using the two antenna method described in MIL-STD-461A. An "in situ" antenna

factor is determined for two identical antennas at specific locations in a given shielded enclosure. To determine antenna factors at an open field site,

the standard antenna approach is used at NBS to calibrate di poles for the

horizontal component of electric field. The standard field method is used for

calibrating monopoles for the vertical component of electric field. Two types

of antennas were used for measurements of antenna factor--vertical monopoles

and horizontal broadband dipoles (biconicals). The antenna factors determined

by NBS open-site calibrations were compared with those determined in the

screenroom by the MIL-STD approach.

The graph given in figure 2A of MIL-STD-461A for a typical biconical

shows variation in antenna factor from a minimum of 7 dB at its resonance dip

to a maximum of 18 dB. The antenna factor curves given in the manufacturer's

manual show a variation from 4 dB (minimum) to 18 dB (maximum). The far-field

antenna factors calibrated at the NBS field site had a total range of 4 dB

(minimum) to 17 dB (maximum). In addition, all of the antenna factor curves

for the two biconicals were within 3 dB of each other over the total frequency

6.3
range of 20-200 MHz.

The layout for measuring the effective gain of two bi conical antennas in

the NBS screenroom involved placing the two antennas on the center line (long

dimension) of the room with both antennas at a height of 1 m above the floor.

The di poles were parallel to each other and horizontally polarized. Two different separation distances were checked, 1 m and 3 m. The effect of screenroom resonances and reflections on the measured antenna factor was much

greater than any of those mentioned in the previous paragraph. The variation

of antenna factor in the screenroom for a separation distance of 3 m was from

54.9 dB (maximum) to -2.3 dB (minimum). The variation for aim separation

distance was 50.1 dB (maximum) to 4.8 dB (minimum). The data also showed

large effects of multipath reflections as a function of antenna placement in

the screenroom. Our data indicate that using open-field-site antenna factors

as permitted in MIL-STD-461A will lead to greater error in field strength

measurements in a screenroom than using antenna factors as measured in-situ.

Measurements of field strength determined from the measured screenroom antenna

factors differ as much as 7 dB from the field measured with a calibrated

transfer probe over the frequency range of 20-200 MHz.

Antenna factors of vertical monopoles are calibrated at NBS by immersing

the receiving monopole in a known field above a groundscreen, about 20 m from

a transmitting monopole. This is done at an outdoor field site which has a 30

X 60 m conducting mesh stretched over a concrete slab. The vertical E com- ponent of the calibrating field is calculated in terms of the base current in

a thin transmitting monopole, at frequencies from 30 kHz to 300 MHz. The

antenna factors measured at the NBS field site were compared with those given

in the manufacturer's manual. The differences were as much as 10 dB at a given frequency. The antenna factors were also measured in the screenroom

using a two-antenna method similar to that described for the biconicals.

Again, two separation distances were used —1 and 3 m. The screenroom values

of antenna factor obtained by the MIL-STD approach differed as much as 64 dB

from that calibrated at the field site.

These measurements indicate that antenna factors determined in a screenroom are not an accurate parameter for subsequent measurement of field

strength. It is concluded that large errors may occur due to room resonances,

multiple reflections, and distortion of fields. The measurements depend on

6.4
the size of the enclosure, location of the test setup in the screenroom, and

position of the antennas. Thus, determination of an in-situ antenna factor in

a screenroom or using antenna factors which were determined in a free-space

environment do not lead to accurate field strength measurements. To establish

accurate EM fields in a screenroom, a small transfer probe should be used to measure the field produced by a radiating antenna.

6,5 Measurement of Broadband Impulsive Signals

The MIL-STD-462 approach for measuring impulse bandwidth, IBW, of a receiver and spectrum amplitude, S(f), of an impulse generator was evaluated

at NBS. The EMI receiver used has a selectable bandwidth which was set in the

1 MHz position for all the data reported here. At each receiver frequency

tested, the gain control was adjusted to obtain a dial indication equal to the

true cw signal level. The impulse generator used for our tests is specified

to produce a spectrum output that is flat within ±1 dB from 60 kHz to 1000

MHz. The IG was set to its maximum output level of 101 dByV/MHz. Photographs

of the receiver response envelopes (video pulses) were used for the MIL-STD

calibration of the EMI receiver. A set of data was taken for at least two

frequencies in each of the eight receiver bands. It was found that the MILSTD measurements of S{f) were within 0.5 dB of the IG nominal output value at

all the frequencies checked below 800 MHz. At higher frequencies the measured

values of S(f) were 0.9 to 2.6 dB above the nominal value.

An impulse generator can also be calibrated by measuring the train of

output pulses with a sampling oscilloscope and taking the Fast Fourier

Transform (FFT) of the average time domain pulse. This approach is used for

the NBS method of calibrating the spectrum amplitude of broadband impulse sources. The S(f) of the IG was calibrated in the frequency range of 50-2400

MHz and these values agreed with the specified IG output within about 1 dB for

all frequencies up to 1 GHz.

Certain "idealized" types of pulses can be described mathematically and

their spectrum amplitude calculated. This approach is used by many manufacturers of IGs, including the IG purchased for these tests. Such "standard"

IGs can be used to calibrate the IBW of a receiver, and are similar to the IG

built into the usual field intensity meter. The IG uses repetitive transient

spikes having a time duration of about 0,5 nanosecond each. The spectrum of

6.5
an IG having such short pulses is fairly flat at frequencies up to about 1 GHz, with an amplitude approximately double the area under the time domain curve of each pulse. It was found that this approximation and the NBS pulse

calibration system using FFT both agreed with the MIL-STD-462 measurements

within about 2 dB at all frequencies checked up to 1 GHz.

6.6
7. Acknowledgments

The authors wish to express their appreciation to all those who

contributed time and interest during this NBS/Army project. We especially

acknowledge the cooperation and assistance of Mr. Gilmore Krenner, Jr., the

contract officer's representative. We are also grateful to Susan Macaskill

and Michael CdeBaca of the NBS staff who accomplished most of the data

gathering and analysis, both in the screenroom facility and at the various

field sites used for obtaining comparison data.

7.1
NB ^ 114A iREv. 2.ec) U.S. DEPT. OF COMM. EIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NBS/TN-1300

2. Performing Organ. Report No 3. Publication Date

October 1986

TITLE AND SUBTITLE

assessment of Error Bounds for Some Typical MIL-STD-461/462 Types of Measurements

5. AUTHOR(S)

J.E. Cruz and E.B. Larsen

6. PERFORMING ORGANIZATION (If joint or other than NBS, see instructions)

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street, City. Stote. ZIP)

U.S. Army Aviation Systems Command

DRSAV-QE

St. Louis, Missouri 63120

10. SUPPLEMENTARY NOTES

QJ Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less factual summary of most significant Information. If document Includes a significant

bibliography or literature survey, mention it here) This report deals with the instrumentation and equations for several systems

used by the U.S. Army for electromagnetic compatibility (EMC) testing and calibrations. Most testing for MIL-STD-461/462 is performed in a shielded enclosure

(screenroom) rather than at an open field site, which leads to uncertainty in the

measurement of emissions from electronic equipment, or the susceptibility of equipment to radiation. Assessment of error bounds by the National Bureau of Standards

(NBS) is covered in this report, and suggestions are given for improving the

measurements.

Four areas of concern were studied as follows: (a) electromagnetic (EM) fields

generated in a parallel-plate transmission line (stripline) , (b) EM fields beneath a single-wire transmission line in a screenroom (long-wire line), (c) determination of

antenna factors for electromagnetic interference (EMI) antennas located in a screenroom, and (d) calibration of EMI receivers to measure broadband impulsive signals.

Most EMC antennas at NBS are calibrated at an open field site or in an anecholc

chamber. This report presents antenna factors determined in a typical screenroom

using the two-antenna method, and comparison with those determined at an open field

site. The video pulse technique prescribed in MIL-STD-462 for calibrating EMI

receivers was also evaluated. Four different methods were tested for comparison

with the MIL-STD approach. They are defined and discussed in this report.

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

antenna factor; electromagnetic compatibility; field strength standards; Impulse

spectrum amplitude; military standards; screenroom measurements

13. AVAILABILITY nn Unlimited

I I For Official Distribution. Do Not Release to NTIS

rx] Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402. (^ Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

108

15. Price

U.S. GOVERNMENT PRINTING OFFICE: 1986- 77 3-66 1/ 65925 USCOMM-DC 8043-P80
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