NBSIR 83-2753

Evaluation of Transient

Measurement Methods in

Gas-Insulated Transmission Lines NAT'L INST. OF STAND & TECH MHHHOMnnWHHHUNnnWMNMnHMMMKIMlIJMMHMHMnMaMMIHnHMHIUMMJHMiyilVna

A 111 Ob OBb^b

U S. DEPARTMENT OF COMMERCE

National Bureau of Standards

National Engineering Laboratory

Center for Electronics and Electrical Engineering

Electrosystems Division

Washington, DC 20234

August 1 983

Prepared for Bonneviile Power Administration

P.O. Box 3621

n ortland, Oregon

QC

100

. U56

33-2753

1933

97208

NBSIR 83-2753

EVALUATION OF TRANSIENT

MEASUREMENT METHODS IN

GAS-INSULATED TRANSMISSION LINES

Ronald H McKnight and Howard K. Schoenwetter

U S. DEPARTMENT OF COMMERCE

National Bureau of Standards

National Engineering Laboratory

Center for Electronics and Electrical Engineering

Electrosystems Division

Washington, DC 20234

August 1 983

Prepared for: Bonneville Power Administration

P.O. Box 3621

Portland, Oregon 97208

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

TABLE OF CONTENTS

Page

LIST OF FIGURES iv

LIST OF TABLES vi i

Abstract 1

1. INTRODUCTION 2

2. EXPERIMENTAL APPROACH 2

2.1 Transient Sheath Potentials 10

2.2 Model for Short Gas-Insulated Conductors 14

3. SENSOR CONSIDERATIONS 18

4. EXPERIMENTAL RESULTS 28

5. LINE DRIVERS AND OPTICAL LINKS FOR FAST

PULSE TRANSMISSION 45

5.1 Line Drivers 45

5.2 Optical Links 58

6. CALIBRATION 60

7. SUMMARY 65

8. ACKNOWLEDGMENTS 66

9.

REFERENCES 67
LIST OF FIGURES

Page

Figure 1. Mercury-wetted relay cable pulser. Driver circuitry

is not shown 4

Figure 2. Coaxial test line for transient measurements.

Line diameter is 0.15 m 5

Figure 3. Schematic view of test line extension and high voltage

spark gap. Extension diameter is ~1 m. Line is

terminated with resistor spider (50 q) 6

Figure 4. Direct pulser output for short charge cable. Output

voltage when driving test line would be ~-20U V 7

Figure 5. Direct pulser output for 305 m of RG58/U. Droop is

due to cable loss 8

Figure 6. Direct pulser output for 46 m of RG214/U. This cable

was used for most measurements reported here 9

Figure 7. High voltage signal produced when line (fig. 2) is charged

to 20 KV and discharged through switch into line extension

(fig. 3). Sensor was plug sensor (fig. 13) 11

Figure 8. Model for short gas-insulated transmission line 12

Figure 9. Equivalent circuit for model shown in figure 3 13

Figure 10. Calculated waveforms for locations 4, 5, and 6 in

figure 8. Source inductance was 1.6 nH. The

assumed source function was a 1 p.u. step 15

Figure 11. Calculations for the line as indicated in figure 10,

except for a larger source inductance (0.16 mH).

Note significant difference in sheath voltage magnitude

between calculations shown in figures 10 and 11 16

Figure 12. Sheath signal measured with plug capacitive sensor

located at location 6, with no charging resistance

between high voltage supply and transmission line

(see text) 17

Figure 13. Plug sensor inserted in "T" section of test line.

Plug diameter is 0.12 m 19

Figure 14. Simplified equivalent circuit for capacitor divider. ... 20

Figure 15. Cross section of disk sensor. Thickness of dielectric

is exaggerated for clarity. Standoff mounting was also used. Measurements were made with and without

shielding skirt 22

TV
LIST OF FIGURES (cont.)

Page

Figure 16. Schematic of model BPA sensor 23

Figure 17. Lumped parameter equivalent circuit used in

theoretical calculations. Various elements were

assigned negligible values during calculations 26

Figure 18. Calculated step response for divider with Ci = 9.4 pF,

C 2 = 95 yF, C 3 = 35.2 pF, R x = 50 n, L8

= 105 nH,

and Lg = 5 nH. All other components were assigned

negligible values (see fig. 17 and table 2) 27

Figure 19. Calculated step for divider with L 9 increased to

25 nH. All other component values are the same as in

figure 18 29

Figure 20. Calculated step response for divider with component

values as in figure 18, except Rll = 50 jj 30

Figure 21. Output voltage from plug sensor mounted in line. Signal

on line was from cable pulser (fig. 1). Charge voltage

was 385 V 31

Figure 22. Output voltage from plug sensor in line. Signal input

was from cable pulser, short cable (fig. 4) 32

Figure 23. Plug sensor output voltage for sensor mounted at end of

extension line. Input signal, figure 6 34

Figure 24. BPA model sensor output voltage for distributed low side

capacitance C2 (see text). Input signal, figure 6 .... 35

Figure 25. BPA model sensor output voltage for single capacitor low

side (see text). Input signal, figure 6 36

Figure 26. BPA model sensor output voltage for distributed low side

capacitance C

2 and 50 q damping resistance between C]_ and

C2

. Input signal, figure 6 37

Figure 27. Voltage output from disk sensor mounted on 0.05 m standoffs, no shielding skirt (see fig. 15) 39

Figure 28. Disk sensor output voltage with shielding skirt

(see fig. 15) 40

Figure 29. Signal from plug sensor mounted at end of extension for

high voltage signal source. Line charged to 20 kV .... 41

Figure 30. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal 42

v
LIST OF FIGURES (cont.)

Page

Figure 31. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal 43

Figure 32. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal 44

Figure 33. Output signal from pulse generator used in evaluating

low side electronics 46

Figure 34. Output signal from probe A (see text) driven by signals

shown in figure 28 47

Figure 35. Output signal from probe A driving 46 m of RG214/U .... 48

Figure 36. Output signal from probe 8 (see text) driven by signals

shown in figure 33. Short output cable 49

Figure 37. Output signal from probe B when driving 46 m of

RG214/U cable 50

Figure 38. Output signal from fast buffer amplifier when driven

directly from pulse generator (fig. 33) 51

Figure 39. Output from fast buffer amplifier driving 46 m of

RG214/U cable 52

Figure 40. Comparisons of measurements of signals on the gas line

for the plug sensor located in-line operated passively

(a) and with active electronics (probe 8) in the low

side (b) 54

Figure 41. Comparisons of passive (a) and active (b) low sides

(see fig. 40) 5b

Figure 42. Comparisons of passive (a) and active (b) low sides

(see fig. 40) 56

Figure 43. Output signal from fast buffer amplifier coupled to

output of plug sensor located in-line 57

Figure 44. Output signals from fast buffer amplifier installed

integral to disk sensor (see fig. 15, text). Signal

on test line was produced by mercury-wetted relay

pulser 59

Figure 45. Step response calculated for plug sensor using circuit

in figure 17 and component values C]_ = 1.01 pF,

C2= 1107 pF, R i = 50 ft, R 2

= 50 ft, and all other

components negligible 62

vi
LIST OF FIGURES (cont.)

Page

Figure 46. Step response calculated for plug sensor using

component values listed in figure 45, except Lq = 1 U H (see fig. 17) 63

LIST OF TABLES

Page

Table 1. Capacitance values and calculated ratios for sensors .... 24

Table 2. Circuit values used in calculations for BPA model

sensor 25

Table 3. In-line plug sensor measurements 60

Table 4. Calculated ac response for plug sensor 64

EVALUATION OF TRANSIENT MEASUREMENT METHODS

IN GAS- INSU LATED TRANSMISSION LINES

Ronald H. McKnight and Howard K. Schoenwetter

Abstract

Capacitive sensors suitable for measuring transients in gas-insulated

transmission lines have been studied in the laboratory. Measurements of

the step response of three different sensors were made with a test line

using both low voltage (200 V) and high voltage (10 kV) signals. Sensor

designs were based on those used in pulse power measurements. The use of

active electronics at the sensor output in the form of fast buffer

amplifiers or commercial FET input probes was investigated as a means of

extending low frequency cutoff. Lumped parameter models were used to

provide theoretical analysis of experimental results.

Key words: capacitor sensors; gas-insulated equipment; high voltage;

transient measurements; transmission lines.
1. INTRODUCTION

The use of gas insulated systems ( G IS ) is steadily increasing

throughout the world. A recent workshop [1] dealt with gas insulated

substations from the user standpoint while a useful bibliography of gas

insulated substations [2] has been prepared by an IEEE working group. One

characteristic of GIS is the production of significant overvoltages during

various switching operations [3-5]. These switching operations can also

result in a transient rise in the external sheath potential of the equipment

[6]. Suitable measuring systems must be used to determine the transient

conditions existing on GIS during switching operations. This is because

of the fast breakdown process associated with the pressurized SFg gas

dielectric used in gas systems. Rise times of breakdown in SFg may be

subnanosecond under certain conditions [7], but are more typically 5- ID ns for restrikes during disconnect operations [8], Electrically, GIS may be

considered as reasonably complex transmission lines, with various surge

impedances and terminations. As a result, fast transients propagate

freely throughout the system with little or no attenuation of high

frequencies. Both experimental observation and modeling using computer

codes for transient analysis have been used to improve understanding of

overvoltage production in GIS. Examples of such research may be found in

the literature [3,5].

Severe requirements are placed on instrumentation to be used in

investigations of transients in (and on) GIS. If information about the

initial transient is required, the system must have nanosecond time

response. In [3], waveforms are shown which contain both MHz and

kHz components. As will be discussed later, the bandwidth of the

system may be requi red to extend down to 60 Hz to facilitate on-site

calibration. A significant transient rise in the potential of the sheath

may be encountered, which can result in unacceptable noise levels and may

damage equipment under severe conditions.

The purpose of the effort reported here was to investigate different

configurations of capacitor sensors which could be used to make measurements

of transients in gas insulated systems. A facility was developed to be

used in investigations of the transient response of capacitor sensors of

various configurations, including near full-scale sensors of the type

installed in the Bonneville Power Administration (BPA) equipment. In addition to experimental work, calculations were made using circuit analysis

codes of lumped parameter equivalent circuits to allow a comparison of

experiment and theory. These and other aspects of the effort are described

in detail in the following sections.

2. EXPERIMENTAL APPROACH

In order to evaluate the performance of capacitive sensors, a suitable

signal source must be provided. Ideally, the step response of the sensor

should be determined since the maximum information about the sensor can be

determined from this type of measurement. In principle, the response of

the sensor to an arbitrary waveform can be determined if the step response

is known [9]. Because the dividers considered here have no discrete high

2
side capacitance, signals cannot be directly applied to the divider. Any

system devised to generate a signal will have electrical characteristics of

its own, which must be known if the di vi der^ response is to be determined.

The use of a gas insulated transmission line for these investigations seems

ideal. The impedance character!’ sti cs of a complex line can be determined

using ref lectometry techniques and the propagation of transients on a line

can be modeled. Step signals can be readi ly generated on a transmission line.

One means of generating a step signal is to use a cable pulser with a mercury-wetted relay. Such a pulser is shown schematically in figure 1. The

charge line is charged to V Q

, then discharged into the gas transmission line,

figure 2. If the line impedances are equal and the gas line is terminated, in

the characterise c impedance, a step will propagate down the line with

magnitude V

0 /2 and step length equal to twice the electrical length of the

charge cable. Because of limitations on the relay, maximum signals are limited to a few hundred volts. This signal level is satisfactory for some types of testing but provides inadequate signal levels if the divider ratios

are 105 or larger. In principle, the pulse can be made as long as desired,

but a practical limit is a few microseconds, due to losses in the charge cable.

A second method for generating a step is to charge the gas line itself

to some voltage and then discharge the line into its characteri Stic impedance.

Since the line can be charged to higher voltage and discharged through a spark gap, much larger signals can be generated in this way than with the

pulser. However, pulse lengths are limited unless a cable extension can

be provided to increase the effective electrical length of the gas line.

In figure 3 is shown a pressurized gas switch along with a large diameter

extension. The gas switch was used to provide high voltage (~ 10 kV) pulses

as described above, while the extension was used to investigate full-scale

sensors of the type found in BPA equipment.

Although the coaxial cables, gas line, and extension are all designed

for a nominal 50 q impedance, there are obvious geometric mismatches at the

input and the line-extension transition. Also, the switch assembly does not

have a constant impedance. Although smoother transitions can be made by using

tapered lines or appropriate designed stepped sections, no attempts were made

to improve the impedance uniformity of the test facility during the work

reported here. The effects of these mismatches resulted in reflections

and a waveform which deviated from a smooth step after approximately 18 ns. These phenomena are discussed in more detail in following sections.

A transient digitizer and associated control unit were used for data

acquisition and manipulation. The digitizer unit has a maximum effective

sampling rate of 100 GHz and was used with a vertical amplifier with a 500 MHz bandwidth and an effective rise time of 0.8 ns. A 100 MHz storage

scope with an effective rise time of 3.5 ns was also used for some measurements. The di gi tizer-cont roll er combination provides hard data

storage as well as the capability for extensive data manipulation and

plotting.

Examples of the output waveform from the cable pulser are shown in

figures 4-6. These data were acquired by using a low charging voltage and

discharging the pulser into the same signal cable used in the divider

measurements. This signal cable was 11.6 m of RG214/u and was used in

all measurements reported here unless noted. Although mercury-wetted

3
H x4 K

x2

x3 L

(b)

Figure 1. (a) Mercury-wetted relay cable pulser. Driver circuitry

is not shown.

(b) Transmission line equivalent circuit. Impedances represent

discontinuities at input to test line.

4
Coaxial test line for transient measurements.

Line diameter is 0.15 m.

Figure 2.
Probe Mixed

Modes

(b)

Figure 3. (a) Schematic view of test line extension and high voltage

spark gap. Extension diameter is ~1 m. Line is

terminated with resistor spider (bu fl).

(b) Transmission line equivalent circuit, showing impedance

mismatch at geometric transition. Propagation involves

mixed modes in extension section.

6
OUTPUT

UOLTAGE

Figure 4. Direct pulser output for short charge cable.

^ Output

voltage when driving test line would be ~^00 V.

7
OUTPUT

UOLTAGE

Figure 5. Direct pulser output for 305 m of RG58/U. Droop is

due to cable loss.

8
OUTPUT

UOLTAGE

Figure 6. Direct pulser output for 46 m of UG214/U. This cable was used for most measurements reported here.

9
relays can produce sub-nanosecond rise time pulses, the pulser configuration

used here had a rise time of approximately 2 ns as shown. Relays are not

available with 50 n geometry, so stray impedances in the pulser assembly

are probably responsible for the structure on the leading edge. The droop

in the signal for the long cable is due to losses in the cable. Although

a faster rise time would be desirable, the signals are adequate for much

of the investigation of the capacitive sensors reported here. In actual

operation, the charge voltages are several hundred volts, but a direct

measurement of the pulser output at these voltages is not possible because

of equipment voltage limitations, i.e., the lack of a suitable high speed,

high voltage attenuator. This is not a serious limitation to the work

presented here.

The high voltage switch was operated at various pressures of dry air

and breakdown characteristics determined for different gap spacings. A gap

spacing of 1.32 mm and a pressure of 0.219 megapascals (30 psi ) allowed a charging voltage of 20 kV. Figure 7 shows the output of a disk configuration

divider (discussed later) for a 20 kV charging voltage. This charging voltage

corresponds to a 10 kV pulse on the line. The rise time is approximately

2 ns, which is comparable to that achieved with the pulser.

The oscillations in the divider output are not associated with the

divider, but appear to be due to the impedance mismatches in the line

discussed earlier. It was also found during the high voltage tests that

there was a transient ground potential increase. This increase was estimated

to be less than 200 V by using a capacitor divider located external to the

test line. A careful series of measurements indicated that reliable data

could be taken using the digitizer and control unit without any special

shielding or noise suppression measures. During the measurements reported

here, the gas transmission line was grounded at several points to a local

ground line. The laboratory floor did not contain a well-defined ground

plane.

2.1 Transient Sheath Potentials

In addition to the transients that are generated inside GIS during

switch operations, it is possible that significant potentials appear on the ground sheath [6], These potentials can cause difficulties in the

measurement process in the field. The possibility of similar transients

exists in the laboratory during operation of the test line in the high

voltage mode. Such transients were indeed observed, but as noted above,

the transient were not sufficiently large to present any measurement

di ff i culties

Preliminary calculations have been made to model the laboratory gas

line during different switching experiments. These calculations, along with

experimental results, are useful in determining possible deleterious effects

on measurements during switching opertions and in aiding investigations of means of accurately measuring sheath transient potentials.

The transmission line used in the calculations is illustrated in

figure 8. Although only one ground strap is indicated in figure 8,

any configuration of different strap locations is allowable. The

equivalent circuit for the model is shown in figure 9.

10
TIME, n*

Figure 7. High voltage signal produced when line (fig. 2) is charged to 20 KV

and discharged through switch into line extension (fig. 3). Sensor

was plug sensor (fig. 13).

11
Outer Conductor

r i t)U re . ;odel for snort Ljas-i nsu 1 ated transmission line,

a breakdown to the outer sheath, or a disconnect

operation in the center conductor.

wnicn nay have

swi tchi n^

12
igure y. Equivalent circuit for model shown in figure 8. Nodes are as indicated in figure 8. Transient voltages resulting from

disconnect operations were calculated for points 4, b and 6

(see figs, lu and 11).

13
2.2 Model for Short Gas-Insulated Conductors

The object of the model is to represent a short gas-insulated conductor

during internal switching or gas breakdown inside the pipe. The model is

digitally simulated by the Electromagnetic Transients Program (EMTP) which

is used as follows. A circuit composed of distributed lines, lumped

impedances, switches, and sources is specified using the circuit nodes for

identification. The EMTP sets up the initial conditions (trapped charge, etc.),

and ti me-i ntegrates the nodal equations. Any node potential or branch current

can be listed and/or plotted. The details of model elements used in the

calculation are discussed below. The conductor and pipe with impedances Z-j and Z

0 , respectively, are considered as two separate transmission lines.

A coaxial transmission line consisting of the conductor and the inner pipe

surface of surge impedance Z-j = 60 In (r-j/r 0 ) where r-j = pipe inner radius

and r 0

= conductor radius is in series with a transmission line consisting of

a cylindrical conductor over the ground plane of surye impedance, Z 0

= 60 In (2h/rz

). Here h is the height of the pipe center above the ground while

r

z is the pipe outer radius.

The source (V) is represented by a 60 Hz sinusoidal voltage and its

associated impedance which, in this case, is assumed to be inductive. The

groundstrap (Lg) is modeled by a lumped inductance of 0.9 yH [6]. The

internal switch (S^) which simulates a disconnect is modeled by a switch

which at its position on the conductor closes at time 0 +

. Trapped charge

•is allowed for. Flashover to the sheath can also be modeled. In this

case S 2 and Zg are modeled by a switch which also closes at time 0. Two

examples of calculations made using EMTP are shown in figures 10 and 11.

Both figures represent a disconnect calculation. The only change in

the two calculations is the value of the source inductance as indicated

in the captions. The order-of-magni tude difference in voltages appearing

on the sheath is indicative of the importance of source impedance in this

calculation. Voltages are shown at three different locations, and the

complex waveforms generated are indicated.

Measurements were made in the laboratory for an experimental

configuration where a gap was placed in the inner conductor of the gas

line, but the outer conductor remained continuous. The gap broke the

coaxial line into two segments of (2/3) L and (1/3) L, where L is the line

length. The longer segment was then charged so that an overvoltage was applied to the gap. Measurements were made of the signal appearing on the

outside sheath using both the plug capacitive divider described below and

a small fast resistor divider. These measurements were made for different

charging resistors placed between the gas line and the high voltage supply.

As the charging resistor was varied from 5 Mq to zero, the observed overvoltage

increased by approximately a factor of 10. These measurements were made

with a single ground strap on the line as in figure 8. However, because

of the presence of supports for the line and a lack of knowledge of the

actual impedance of the high voltage source, it is not expected that there

would be quantitative agreement between theory and experiment. An example

of the signals observed is shown in figure 12. These were taken at a location correspondi ng to location 6 in figure 8 using the plug capacitive

sensor, located beneath the transmission line at a distance of approximately

10 cm.

14
Sheath

Voltage,

p.u.

0 20 40 60 80 100 120 140 160 180 200

Time, ns

Figure 10. Calculated waveforms for locations 4, 5, and 6 in

figure 8. Source inductance was 1.6 nH. The

assumed source function was a 1 p.u. step.

15
Sheath

Voltage,

p.u.

0.90

0.68

0 20 40 60 80 100 120 140 160 180 200

Time, ns

Figure 11. Calculations for the line as indicated in figure 10,

except for a larger source inductance (0.16 mH).

Note significant difference in sheath voltage magnitude

between calculations shown in figures 10 and 11.

16
Figure 12. Sheath signal measured with plug capacitive sensor located at

location 6, with no charging resistance between high voltage

supply and transmission line (see text).

17
These measurements indicated that for a charging resistance of 5 Mft and a charging voltage of 10 kV, transient signals on the sheath were less

than 200 V. A series of experiments were done which indicated that data

could be taken using the digitizer and controls even when the line was operated in a high voltage mode, when the extension was in place.

The results and calculations for sheath potential rise discussed

above are preliminary and do not represent an extensive effort. They are

a useful introduction to the use of theory and experiment to investigate

transient phenomena generated on the sheath of a gas transmission line in

the laboratory.

3. SENSOR CONSIDERATIONS

Although GIS are usually equipped with precision voltage and current

measuring devices, these are not useful for transient measurements because

of restricted bandwidth. An example of instrumentation designed specifically

for GIS, but not for transient measurements, is found in [10].

Conventional impulse dividers have been used in certain measurements [3,6],

but cannot be used to detect transient voltages internal to the gas equipment.

They are also limited in high frequency response. Fast compensated resistor

dividers have been used in pulse power systems [11], but in this case there is no steady-state voltage and low impedances can be tolerated. Simple capacitor

sensors have been used in short pulse high voltage applications for many

years [12,13] and appear to be ideally suited, with suitable modifications,

for use in GIS. High voltage capacitor dividers for freestanding use have

been described in the literature [14] and in measurement textbooks [16,16].

One configuration of a divider suitable for use in gas insulated equipment

is shown in figure 13, while a simple equivalent circuit is shown in figure 14.

The advantages and disadvantages of this (and similar) dividers may be seen from figure 14 and the calculated step response,

V (t) = l r 2_\ exp (-t/ T )

» \Qi u)

where the time constant x of the circuit is given by

t = (Ci+C2 )(Rl+R2) • (2)

It is generally recognized that these capacitor dividers which are integral

to the system are capable of subnanosecond response and this has been

shown in other applications [7,8]. The bandwidth problem lies in the low

frequency response. If the transmission line is terminated in its

characteristic impedance to avoid reflections at the measuring device, then

the practical lower cutoff frequency can be high. For example, the calculated

low frequency cutoff with Ri and R 2 equal to 50 ohms is

f co = 1 / ( 2-rrx ) - 1.5 to 0.15 MHz (3)

for typical C 2 values of 1 to 10 nF.

This analysis is based on the simple treatment of the circuit as a high

pass filter. This cutoff situation can be improved by increasing R

2

. In

18
Inner Conductor

Figure 13. Plug sensor inserted in "T" section of test line.

Plug diameter is 0.12 m.

19
o—

v^t)

o

a/W

Q~

Figure 14. Simplified equivalent circuit for capacitor divider.

20
a later section, consideration is given to the use of active electronic

circuitry in the low divider output side to extend the cutoff frequency to below

60Hz. A second problem is associated with the fact that Cj. is not a discrete

component, but represents the coupling capacitance between the sensor and

the high voltage electrode. This means that the divider cannot be evaluated

as a stand-alone unit. The step response of the divider and its attendant

measuring system can be studied in the laboratory, but not with the identical

C]_ that exists in the actual system. As a result, the final calibration

of the divider must use methods suitable for an in-place calibration. This

is a non-trivial problem. With regard to calibration, it should be made

clear that the achievable accuracies associated with these measurements

will not be comparable to those of potential transformers , although they

should be adequate for anticipated requirements.

Other special requirements may be placed on the design of a suitable

capacitor divider. Voltage limitations on low side components may place

requirements on the (4/C2 ratio. If active components and optical links are to be used [17], then overall system integration must be considered.

Physical constraints may dictate the geometry of the divider. Despite all

these special conditions, it is possible to learn a significant amount by

doing laboratory investigations of model or full-scale sensors.

A series of measurements was made using three different divider

configurations. One of these, the "plug" sensor, is shown in figure 13.

In the plug sensor, there is almost complete shielding except for the

portion of the center element exposed to the high voltage electrode. The

low side capacitor C2

is an ai r-di elect ri c coaxial capacitor. Three-termi nal

capacitor measurements were made of (4 and C2 for this sensor installed

in the tee of the transmission line (see fig. 2). The values obtained

were 1.012 pF and 1.107 nF, respectively.

A second configuration is shown in figure 15. Here C2 is a folded

parallel plate capacitor. The dielectric was 10 mm thick polycarbonate.

This material was chosen because of availability at the time of divider

construction. The ratio of the divider can be changed by varying the size

of the opening in the top plate, which affects both (4 and C2

. Other

dielectric materials may be used as well. The thickness of the brass

plates making up the unit was 0.79 mm. This is significantly greater

than the 1 GHz skin depth.

The third divider (the BPA divider) is shown schematically in

figure 16. The sensor, which is 0.4 m (16 in) in diameter, was intended

to simulate the disk sensors installed in BPA equipment. The spacing above

the sheath was chosen to be approximately that for the actual sensor. A

nylon feedthrough carried the signal from the disk through the enclosure

wall to a small shielded box in which C

2 was mounted.

Two different configurations for C 2 were used in these investigations.

The purpose of changing C 2 was to vary the inductance associated with the

low side capacitance. To produce a minimum inductance configuration,

10, 0.01 yF, miniature ceramic capacitors were soldered between two brass

plates. The capacitors used were designed for microwave circuitry.

21
Shield

Figure 15. Cross section of disk sensor. Thickness of dielectric is

exaggerated for clarity. Standoff mounting was also used.

Measurements were made with and without shielding skirt.

22
Center

Electrode

V*-

_L

T

ci Sensor

Figure 16. Schematic of model BPA sensor.

23
This assembly was mounted on the feedthrough and one plate was grounded

directly to the shield. A second, more inductive, low side consisted of a single 93 nF disc ceramic capacitor connected between the feedthrough and

the box.

Dividers with coaxial geometry such as those described in [7,18] were

not considered. This was because thei r installation in existing equipment

requires significant disassembly, whereas the plug or disk sensors can be

readily installed on existing flanges. They do represent a viable alternative

to disc-type sensors for new equipment design.

Table 1. Capacitance values and calculated ratios for sensors

Cl

PLUG SENSOR - in line 1.012pF a

PLUG SENSOR - end 0.46pFb

DISK SENSOR - 2" spacer 0.4pFb

BPA MODEL SENSOR 9.4pF*>

a measured

^calculated

Calculated for Rj = R

2 = 50 n (see fig. 14)

C2 Ratio 0

1107 pF a 4.57 x 10-4

1107 pF a 2.08 x 20-4

1340pF a 1.49 x 10-4

96300pF a 4.88 x 10“

The plug sensor could be mounted either in the tee of the transmission

line (fig. 2) or in the end of the extension (fig. 3). The disk and BPA

dividers could only be mounted at the end of the line. Table 1 summarizes the

measured or calculated capacitances associated with the three dividers, and

their ratios based on eq (1). The results for the three sensors will be

presented as they occurred chronologically, since the development of the disk

sensor was based to some extent on results obtained from the other two sensors.

A more realistic equivalent circuit for a capacitor or divider than

that shown in figure 14 would be one in which various stray capacitances and

inductances are included. The inductances are particularly important

because of the large impedances resulting from high frequency signals. A more fundamental question concerns the validity of using lumped parameter

equivalent circuits. Skin effect phenomena at high frequencies cause signal

propagation along, rather than through, conductors so that a transmission

line approach may be more relevant. Wave propagation on the divider itself

may result in oscillations which are not predicted from the lumped parameter

model. The frequency of these oscillations would be geometry dependent.

This means that evaluations of various divider configurations must be made

with near-ful 1 -scale configurations. Oscillations of scaled models, which

are beyond the range of measuring instruments and therefore unobserved,

may become dominant for full-scale devices.

24
The simple circuit of figure 14 cannot be expected to model adequately

real capacitor dividers. A more realistic circuit is shown in figure 17.

Here various stray elements have been included. Other circuits could be

proposed but the salient features of the experimental results can be explained

qualitatively using this model. The complicated ladder network around the

low-side capacitor C 2 was not used in the present calculations but has

been retained for possible future work. The capacitance to ground C3

is an important part of the BPA sensor. It results from the location of the

sensor above the ground plane. For the plug and disk sensors, this stray

capacitance is negligible.

The main capacitance elements Ci, C2 , and C3 can be measured or calculated.

Resistances Rn, Ri, and R 2 are discrete components. Stray inductances L3

and Lg cannot be measured but their composite value was deduced by comparing

theory and experiment. This was done by using known (calculated) values for

Ci, C

2

, C

3 , Ri, R2 , and Rn, and varying Lg until the step response ringing

approximately matched the experimental value of 73 MHz. Then Lq and Lg were

varied until both ringing frequency and divider ratio approximately matched

experimental results. The goal of this exercise was to obtain an equivalent

circuit which provided reasonable agreement with experimental results.

Circuit values obtained in this way are given in table 2. The inductance

values are small, but their effect is dramatic. The results of calculations

are shown in a series of figures beginning with figure 18.

Table 2. Circuit values used in calculations for BPA model sensor 3

Cl

C2

C3

I b

l8

l9

R 1

R 2

a Refer to figure 17. Rg, L4 , Lg

negligible values.

bThe total inductance for Lg plu:

cThe 5 q, value was required to r the 50 q value was used to calci

29.4 pF

95 nF

35.2 pF

105 nF

5 nF

50 n

50 n

5 fl, 50 n

L

7

, L^q , C

5 , C5 were assigned

Lg was 110 nF.

produce the observed damping, while

late the overdamped response.

25
Figure 17. Lumped parameter equivalent circuit used in theoretical

calculations. Various elements were assigned negligible

values during calculations.

26
STEP

RESPONSE

TINE, ns

Figure 18. Calculated step response for divider with Ci = 9.4 pF,

C 2 = 95 U F, C 3

= 35.2 pF, R \ = 50 q, Lg = 105 nH,

and Lg = 5 nH. All other components were assigned

negligible values (see fig. 17 and table 2)

27
In figures 18 and 19, results for two different values of Lg are shown. The

ringing frequency changes are small but the amplitude of the step response

changes significantly. When = 50 n, a damped response is obtained, as

is shown in figure 20. This should be compared to figure 26 below. Note that

the 600 MHz signal seen in the lab (fig. 24) is not present in the step response.

From these calculations, it is clear that very small inductances in the

low side cause significant deviations from capacitive response. This is not

a new result and the importance of good low-side capacitor design for use in

impulse dividers is discussed in the literature [14,25]. Tne absence of

observable oscillations which can be assigned to the dividers indicates that

there is very little inductance associated with either the plug or disk

dividers. This is perhaps to be expected since the low sides in each case

are short transmission lines. For the plug sensor, the calculated inductance

per meter is 0.8 nH, which corresponds to a total L of 0.06 nH. A meaningful

similar calculation for the disk cannot be readily made. Because the

outputs of these low-side transmission lines are not terminated in their

characteristic impedance, there could be ringing. However, because of the

small dimensions involved, this ringing would be in the GHz or higher range.

Despite the list of problem areas, fast capacitive dividers have

been widely used to measure pulses of short duration, particularly in

pulse power systems. A recent workshop highlighted measurements of

electrical quantities in pulse power systems, and several papers presented

at the workshop addressed various aspects of these capacitive sensors [18-21],

Other examples of applications are found in the literature [22-24].

4. EXPERIMENTAL RESULTS

The plug sensor was chosen for use as a reference sensor because earlier

reports [13,18] indicated its usefulness for short pulses, as well as the

fact that it could be located both in-line and at the end of the transmission

line. Figures 21(a-c) show output signals from the plug sensor for an in-line

location, where the pulse on the transmission line was generated by the

relay pulser and corresponds to figure 6. (In fig. 21 and others following,

the same signal is shown on different time scales.) Unless noted, all

experimental results are direct outputs from the sensor for a given input

signal. The rise time of the signal is not significantly changed from that

of the input pulse, and the time constant of the output signal is clearly

evident in figure 21(c). The disturbances on the signal at approximately

18 ns after the start of the pulse appear to be due to reflections from the

geometric transition at the line extension. There is no obvious oscillation

or overshoot associated with the divider itself, at least within the bandwidth

of the measuring system. It should be noted that the pulser output was very reproducible so that the consecutive records were nearly identical.

The response of the divider to a short pulse is indicated in figure 22.

In all of the records taken with this sensor, there was no indication that

divider response deviated significantly from that predicted by eq (1).

The observed waveforms assume a different character when the sensor is

mounted at the end of the line. Data were acquired for both a flush

mounting, where the face of the divider was flush with the end and for a configuration where the divider assembly was inserted into the line up to

the mounting flange. A limited amount of data was acquired for the inserted

confi gurati on and, except for a change in ratio (due to the larger C^),

28
-•.•2

0 23 38

TIME, tm

Figure 19. Calculated step for divider with Lg increased to 25 nH. All

other component values are the same as in figure 18.

29
cnUi

Figure 20. Calculated step response for divider with component values as in figure 18, except Rll = 50ft.

30
260 mV

-208 * ' « *« . • 16 29

(b)

Figure 21. CXjtput voltage from plug sensor mounted in line. Signal on line was from cable pulser (fig. 1). Charge voltage was 385 V.

31
Figure 22. Output voltage from plug sensor in line. Signal input was from cable pulser, short cable (fig. 4).

32
the waveforms were identical. Only results for the flush mount are shown here. Figures 23(a-c) show the waveform present at the end of the

transition section. Here the waveform exhibits an initial overshoot with

a number of subsequent oscillations. A signal with this signature is

characteristic of the response of many dividers to a voltage step. However,

what is observed here for all practical purposes is the actual waveform

exciting the probe. If the divider did have an oscillatory response, it

would have been present for the in-line signal as well (fig. 21). The

changes in waveform from the in-line to the end location are probably due

to the construction of the termination, which is a spider arrangement with

six legs, and the propagation of the wavefront in the extension. The

overshoot in the signal observed at the end of the line is seen on different

sensors and appears to be due to a complicated propagation mode in the

large extension, which results in some voltage enhancement. This overshoot

is also observed for high voltage pulses (see fig. 7). The results shown

in figures 21-23 and those below are typical of many measurements taken

for different charge cables, charging voltages, etc.

The BPA model sensor was mounted in the end of the extension (fig. 3),

and the response was determined for different signals on the line, with

different low side configurations. A large number of records were obtained,

and those shown here have been selected to illustrate specific points.

The signals shown in figures 24 (a-c ) clearly indicate oscillations at two

frequencies (600 and 73 MHz). Although not obvious from the figures, the

magnitude of the signal is approximately 75 times greater than that

predicted by using the calculated ratio (table 1), and the measured charge

voltage. A similar result was reported in [14]. The data shown here are for a minimum inductance low side capacitance. Figures 25(a-c) were obtained

with a single C2 , with greater inductance. The oscillatory character is not

changed significantly, but the output voltage is now some 360 times the

predicted value. These waveforms should be compared to those for the plug

sensor (fig. 23).

These responses show that the BPA sensor as configured in the laboratory

is not useful for measurements. An equivalent circuit described earlier

was developed which was used to explain the observations. The use of both

theoretical calculations and experimental results in understanding the

performance of various dividers is a critical element in developing optimum

divider configurations. The results of these calculations, which were discussed in detail earlier, indicated that inductance in the circuit below

Ci was responsible for the 73 MHz oscillation and the large signals. The

source of the 600 MHz oscillations has not been determined, but measurements

made with the disk sensor suggest that it may be an oscillation in the line

extension which couples strongly to dividers located above the ground plane.

This 600 MHz signal is barely discernable in the output of the plug sensor. Changing the spacing of the BPA model sensor plate from 10 to 5 cm above

the ground plane did not modify the time dependence of the output in any

meaningful way.

To investigate the effect of damping in the divider, a 50 n resistor

was inserted between the sensor plate and the feedthrough, inside the extension.

Figures 26(a,b) show the waveform resulting from this modification. The

effect of the damping is obvious, but the overshoot is still significant,

and the divider is still unacceptable.

33
40 *U

-40 nU • * * * * * * 0 19 20

TINE, m

TIHEi n*

|

(b

)

(c)

Figure 23. Plug sensor output voltage for sensor mounted at end of

extension line. Input signal, fig. 6.

34
s

• *u

3O

(a)

-2088 »U ‘ * ' • 18 28

TIME, n*

TIME, ns

(b)

(c)

Figure 24. BPA model sensor output voltage for distributed low side

capacitance C 2 (see text). Input signal, fig. 6.

35
J HOLTS

e

=> a OOLTS 3-fkO

-4 UOLTS » ' * • tO 20

TIME, t\s

(a)

(b)

(c)

Figure 25. BPA model sensor output voltage for single capacitor low side

(see text). Input signal, fig. 6.

36
808 nU

-800 wV L

23

TIME* ns

58

Figure 26. BPA model sensor output voltage for distributed low side

capacitance C 2 and 50ft damping resistance between and C 2

. Input signal , fig. 6.

37
Based on these results, the disk divider was designed and constructed.

This design, with suitable engineering, could replace the BPA sensor with

a minimum of modification to existing equipment. It was also used in

determining the origin of the 73 MHz oscillation seen in figure 24.

The disk divider could be mounted directly on the feedthrough used with

the BPA sensor, or on standoffs on a mounting plate in which the signal

connection was made to a type N bulkhead connector. As with the previous

dividers, only records illustrating significant results are presented

here. For signals shown in figures 27(a-c), the divider was mounted on 5 cm standoffs, and R]_ (fig. 14) was connected between the divider and the

output connector inside the line. A coaxial grounded shield was positioned

around R]_. When compared with figure 23 and figure 24, the difference is

striking. The 73 MHz signal is gone, as expected, but the 600 MHz signal

is still present. The output voltages are significantly lower for the

disk sensor, although still larger than predicted.

Both the unshielded disk divider and the BPA sensor detected the

600 MHz signal, although their geometries were dissimilar. When the

underside of the disk sensor, which is at dc ground potential, was shielded

with a skirt, as indicated in figure 15, the signal was substantially

reduced as is shown in figure 28. Experiments with improved shielding

indicated that the oscillatory component of the signal seen in figure 28

could virtually be eliminated.

All of the results described above were obtained for low voltage,

cable pulser-generated signals and no active electronics. Signals were

also generated using the gas line and pressurized switch. In this mode,

the system operated as a simple cable pulser which, in principle, would

generate a 40 ns square wave of 10 k V amplitude for a 20 kV charging voltage.

Charging voltages were measured with an electrostatic voltmeter. Examples of

the waveform obtained are shown in figures 29(a-c), which were obtained

with the plug divider. The overshoot at pulse onset and the oscillations

observed on the wavetop are seen on the pulser-generated signals, indicating

that they are real and associated with wave propagation through the switch

assembly and in the extension. The signal rise time is excellent, which

implies a low inductance in the switch itself. For these measurements,

switch gas pressure was 0.219 MPa.

The outputs from plug and optimally shielded disk dividers can be

compared directly, as in figures 30, 31, and 32, where the data shown were acquired for both pulser and high voltage signals. For the same input

signal, the outputs are virtually identical. This is an important result,

since it indicates that two sensors of significantly different geometries

respond in the same way to the same signal. There is no significant distortion

of the signal other than that predicted by eq (1). It also implies that

the complex waveforms shown in figures 30-32 are indeed those exciting the

sensors, and not a result of divider response.

In summary, experimental results show that the BPA sensor cannot

be used to provide meaningful data regarding transients in gas-insulated

equipment. Two alternative divider geometries have excellent transient

characteri sties , but both show the expected RC shaping which, without

modification, limits their usefulness to transients with nanosecond duration.

38
200 *V

la

o

<s£

= • hM

TIME* ns

(a)

(b)

(c)

Figure 27. Voltage output from disk sensor mounted on 0.05 m standoffs,

no shielding skirt (see fig. 15).

39
80 **U

!• TIME, n* M

TIHE, ns

(b)

(c)

Figure 28. Disk sensor output voltage with shielding skirt (see fig. 15).

40
468 rU

=> 8 PHI -

9

a.9

(a)

-a ea *-* 1 • « 19 29

TIME, ns

(b)

(c)

Figure 29. Signal from plug sensor

voltage signal source. mounted at end of extension for high

Line charged to 20 kV.

41
400 *U

Ui

vj

cc§ > mV

-468 *U l

0 j— i 18 TIME, ns28

(a)

Figure 30. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal.

42
48a HJ

-488 «U

(a)

25

TIHE, Tis

58

(b)

Figure 31. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal.

43
400 nil

TIHE. ns

(a)

(b)

Figure 32. Comparison of output voltage from plug sensor (a) and

disk sensor (b) for high voltage signal.

44
5. LINE DRIVERS AND OPTICAL LINKS FOR FAST PULSE TRANSMISSION

5.1 Li ne Dri vers

As was discussed in earlier sections, the low frequency response of the

capacitive dividers can be extended by orders of magnitude by using high

input impedance active electronics at the output of the divider. These

electronic devices, in turn, will be characteri zed by some bandwidth and

rise time so that the system response will be a composite. Care must be

exercised in mating the active probes to the capacitive sensor to ensure

that no spurious responses are produced.

Three different active systems were evaluated in the laboratory and two

of these were used with sensors described earlier. Two wide-band active probes

and a fast buffer amplifier were tested for possible use as a fast-pulse

coaxial cable driver. (As mentioned later, these units may also be suitable

for modulating the transmitter in fiber optic telemetry links.) These

devices were first tested with input voltage steps obtained from a commercial

pulse generator. Measurement and storage of all waveforms from the above

equipment were made with the digitizer system described in earlier sections.

Figures 33(a-c) show a voltage step from the pulse generator applied

directly to the digitizer using three different sweep speeds. The specified

rise times (10-90%) of the digitizer system used and pulse generator are 0.8 and 3.5 ns, respectively.

Figures 34(a-c) show the effect on these waveforms of inserting a commercial

FET probe (probe A) between the pulse generator and digitizer. This probe

has unity gain, 100 kft input impedance, a rise time of approximately 0.4 ns,

and is capable of driving a 50 q, load with signal levels up to ±600 mV.

A long signal cable (150 ft of RG214/U) was also used between the probe

output and the digitizer. The attenuation of high frequencies in the

cable is indicated in figures 35(a,b) which show the decrease in rise time

caused by inserting 150 ft of RG214/U coaxial cable between the probe

output and digitizer. The attenuation of this cable at 50, 100, and 200 MHz

is approximately 2, 3, and 4.5 dB, respectively. While not a problem in

the present work, the effects of long cables on fast signals can be important.

Computer corrections can be made for this signal distortion [26].

Figures 36(a-c) show the system response when a different commercial

probe is inserted between the pulse generator and digitizer. The second

probe (probe B) has a 10/1 attenuation, a 10 Mn input impedance, and a 0.7 ns rise time. Figure 37 shows the effect of inserting the long RG214/U

cable between the probe output and digitizer. In figures 36(a-c), the

voltage scale reflects the input voltage levels to the probe.

An alternative to using commercial FET input probes is to use a fast buffer

amplifier. These amplifiers have bandwidths greater than 100 MHz, slew rates

up to 6000 V/yS, and have been used in fast pulse measurements [27] . These

amplifiers were examined for their usefulness in the present investigation.^-

Figures 38(a-c) and 39(a,b) show the system responses when a buffer

amplifier is substituted for the probe. For these measurements, a 3.74:1

-'•Special circuit boards based on this design were kindly supplied by

Glen McDuff, Texas Tech University, Lubbock, Texas.

45
OUTPUT

UOtTAGE

OUTPUT

UOLTAGE

OUTPUT

UOLTAGE

209 «U

(a)

TIME, n*

TIME, r>s

Figure 23. Output signal from pulse generator used in evaluating

low side electronics.

46
200 *U

-200 rU * * ' • 9 tO 29

TIME, ns

Figure 34. CXjtput signal from probe A (see text) driven by signals

shown in figure 28.

47
OUTPUT

UOLTAGE

OUTPUT

UOLTAGE

209 MJ

0 nU

(a)

-200 wU *- 0 23 TIME i ns

30

(b)

Figure 35. Output signal from probe A driving 46 m of RG214/U.

48
see nu

-680 mi)

(a)

18 TIME, ns

28

(b)

(c

Figure 36. Output signal from probe B (see text) driven by signals shown

in figure 33. Short output cable.

49
Figure 37. Output signal from probe B when driving 46 m of RG214/U cable.

50
•• MJ

§

-ee *o

(a)

it TIME, ns

20

(b)

TIME, n*

uiu

<x

(c)

Figure 38. CXjtput signal fror. fast buffer amplifier v/hen driven directly

from pulse generator (fig. 33).

51
88 nU

uiu

<xo= I (tU

(a)

-00

TIME, ns

(b)

Figure 39. Output from fast buffer amplifier driving 46 m of RG214/U cable.

62
attenuator (13.65 Mn impedance, shunted with 2.2 pF capacitance) was added

to the input of the amplifier to minimize the overshoot characteri sti c of

this amplifier. Since the input capacitance of amplifiers with series

feedback usually increases with frequency in the upper range of their frequencies,

overshoot in these amplifiers can often be decreased by suitable choice of

capacitor values in an input attenuator. A capacitor ratio of ~3.74 was the

smallest ratio that could be used to minimize overshoot in the amplifier

employed in these measurements. Hence, the choice of 3.74 for the attenuation.

Comparisons of waveforms shown in figures 29(a-b) and 31(a-c) with those

in figures 28(a-c) show that both commercial probes are suitable as cable

drivers, subject to the output signal level constraint of ±600 mV on probe A.

Depending upon the input signal level, the 10:1 attenuation of the second

probe may be undesirable. Also, the relatively low input impedance (100 k n)

of the first makes this probe less useful as it limits the low frequency

response of the composite systems.

Combining the rise times of the pulse generator and buffer amplifier in

root-sum-of-squares fashion yields a rise time of about 8 ns for the buffer.

As indicated in later measurements, when the amplifier was used without an input attenuator, the rise time decreased with increased overshoot.

Tests were next conducted using the model gas-insulated transmission

line illustrated in figure 2. The signal was produced using the mercurywetted pulser described earlier. The voltage of the inner conductor was measured using the NBS plug sensor shown in figure 13. Figure 40(a) shows

the result of a measurement of the sensor output, using probe B, while

figure 40(b) shows the sensor output connected in a passive mode. Because

of the divider action of R\ and R 2 (fig. 14) there is a factor of 2 difference in the signal levels between figures 35(a) and 35(b).

A comparison of figures 40(a) and 40(b) indicates that the probe reproduces

perturbations in the model line reasonably well. Similar results were obtained

for probe A. Figures 41(a,b) and 42(a,b) show the plug sensor output waveform,

employing the probe B for the sweep speeds used in previous measurements.

Figures 43(a,b) show the response of the buffer amplifier when fed by

the plug sensor, using the same connecting cable used in all of the previous

measurements. It is believed that the considerable amount of ringing was largely caused by the long lead path between the sensor output (starting

at the top of R]_, fig. 13) and the input terminal of the amplifier. This

path included an adapter for connecting between N- and BNC-type connectors.

To evaluate the buffer amplifier when placed integral to a capacitor

sensor, the amplifier circuit board was mounted in the disk sensor at the

location of R]_ (fig. 15). The sensor output electrode (Cj_) was connected

directly to the buffer input with a very short lead. Also, since the

signal level was only 22 mV (44 mV open-circuit), the 3.7:1 attenuator was removed and a 10 Mq resistor connected from input to ground. The ground

plane of the circuit board was connected to the sensor ground electrode

and the connector ground with pairs of short copper straps. The ±15 V power

supply voltages were each filtered on the amplifier circuit board by two U.l y

F ceramic capacitors and a 4.7 yF solid tantalum capacitor. Further filtering

of these voltages was provided by feeding the power supply leads into the

sensor package by 0.001 U

F ceramic feedthrough capacitors and a ferrite bead.

53
80 nil

-ee • * * • • 0 10 20

TIME i us

(b)

Figure 40. Comparisons of measurements of signals on the gas line

for the plug sensor located in-line operated passively

(a) and with active electronics (probe B) in the low

side (b)

54
OUTPUT

UOLTAGE

OUTPUT

UOLTAGE

89 «U

9 mU

(a)

-88 "U

30

TIHE. ns

100

Figure 41. Comparisons of passive (a) and active (b) low sides

(see fig. 40).

55
200 Mi

Ui

<z

d 9 nU

-200 hU l

9 250

TIHE* ns

309

(a)

(b)

Figure 42. Comparisons of passive (a) and active (b) low sides

(see fig. 40).

56
80 wU

Uio

«x§ • *u

(a)

-68 *<J ' ‘ ' • * ' 1 • II 29

TIME, nj

(b)

Figure 43. Output signal from fast buffer amplifier coupled to output

of plug sensor located in line.

57
Figures 44(a) and 44(b) show the output waveform from the buffered disc

sensor. Based on previous measurements made on the unattenuated buffer

amplifier (not shown), approximately two-thirds of the overshoot shown in

those figures is attributable to the amplifier. When compared with figure 28,

the decrease in rise time and the loss of high frequency detail is obvious.

5.2 Optical Links

The chief advantages of the fiber optic telemetry links over coaxial

cable links are their immunity from electromagnetic interference and the

small loss of signal strength and bandwidth when the signal is transmitted

long distances. Fiber optic telemetry links may generally be classed as either digital or analog systems. This discussion is limited to analog

systems employing an amplitude modulated light-emitting diode (LED) or a laser diode as the transmitter element. The detector used at the receiving

end of these systems is usually a PIN diode or an avalanche photo diode (APD).

Since the detector is a current source, a transimpedance amplifier (currentto-voltage converter) must be used with the detector to provide a useful

output. In most systems, the detector-ampl i f i er bandwidth determines the

system bandwidth. The detector-amplifier bandwidth can be maximized by

using a low gain amplifier. This is feasible if a large optical signal is

received, or if the detector sensitivity is large. Avalanche diodes have

one to two orders of magnitude greater sensitivity than PIN diodes and,

therefore, facilitate the design of wideband receivers. Their disadvantages

are a gain-dependence on temperature, high noise, and high cost.

The modulator of the transmitting element (LED or laser diode) must

provide peak-to-peak drive currents ranging up to at least ±10 nA, usually

capacitively coupled to a 50 si load. The modulator bandwidth should be

sufficiently large to have minimal affect on the system bandwidth. Continuous

laser diodes are favored over LEDs for wideband applications because of

their higher power output and faster rise times. The narrow spectral

linewidth of lasers (2-4 nm, as compared to approximately 50 nm for LEDs)

is important to minimize fiber material dispersion if the length of the

fiber cable is larger than about 100 m. The disadvantages of laser

diodes, relative to LEDs, are their current dependence on temperature,

poor linearity, and higher cost. The first two disadvantages can be minimized

by careful regulation of the base-value current (an additional cost) and

utilizing a small percent (depth) of modulation (<50%).

Fiber optic telemetry links (complete systems) range in price from

approximately $500 for a 25 MHz system to $4000 for a 200 MHz system. Prices

up to $20,000 are quoted for the 200 to 500 MHz range.

It is not known whether the modulator (or input amplifier) of any of the

wideband commercial transmitters can be directly interfaced with a capacitordivider type of sensor, such as the NBS disc sensor. Therefore, it may be

necessary to employ a special amplifier for interfacing between the sensor

and the transmitter. Possibly, one of the probes or the amplifier, evaluated

for coaxial cable driver use, could be used. The telemetry system bandwidth

requi rements , along with the signal level requirement of the waveform measuring

and recording equipment, will determine the receiver requirements.

58
48 nU

«uo

° t nU

(a)

-40 *u » - ' ' > ' > ‘ • 10 20

, TIME, ns

(b)

Figure 44. Output signals from fast buffer amplifier installed integral

to disk sensor (see fig. 15, text). Signal on test line was produced by mercury-wetted relay pulser.

59
6. CALIBRATION

Although the transient response of the dividers can be studied in the

laboratory, their ultimate application is in field installations. The high

voltage capacitor (fig. 17) does not exist as a discrete component and

its value depends on the geometrical relationship between the divider assembly

and the high voltage electrode. In a properly designed divider, this should

be the only difference between laboratory and field installation. In the

laboratory, a three-termi nal bridge measurement may be used to determine

Ci . Such a measurement may only be practical in the field during installation

In either case, the value of Cl is small, on the order of 1 pF. For the

plug sensor, bridge measurements were made for Ci and C

2 (fig. 17) for the

in-line location. The ratio in table 1 is based on these capacitance

measurements. The experimental ratio determined from the step response

for a number of different measurements is shown in table 3. There is some disagreement between the two. However, there are systematic errors associated with the experimental numbers. One of these errors has to do

with the mismatch at the transition between the pulser output and the gas line

This mismatch will result in a reflection which will reduce the transmitted

pulse below its theoretical value. This error is in a direction consistent

with the observed discrepancy. There are small losses in the signal cable.

No individual calibrations were made of the digitizer or the FET probe.

As a result, the uncertainty of the experimental values of the ratio may

be as large as ±10%. With this consideration and those stated above,

the agreement is reasonable.

One procedure proposed to allow determination of Ci in the field is to

compare the divider output at 60 Hz with a calibrated voltage sensor, say a potential transformer. In principle, everything about the divider system

except C} is known, once it has been evaluated in the laboratory. This

procedure has some severe problems, as can be shown by a theoretical analysis.

Table 3. In-line plug sensor measurements

Measurements Measured ratio 3 Calculated ratio*3

(1) 4.02 X 10"A 4.57 X 10 ~ A

(2) 4.20 X io-a 4.57 X 10 " A

(3) 9.02 X 10 -4C 9.14 X 10 "A

(4) 8.91 X 10 "AC 9.14 X 10 " A

(5) 4.09 X 10 "A 4.57 X 10 " A

(6) 4.17 X 10 "A 4.57 X 10 " A

a Based on actual voltage measurements.

^Based on measurements of circuit parameters.

Measurements (3) and (4) were made with the FET probe. As noted in the

text, this changes the effective divider ratio by a factor of 2.

60
Using the equivalent circuit of figure 17, frequency response was calculated

using component values for the plug sensor. Both low impedance and high

impedance values were used for R1 and R2 to reflect the use of passive or active low side signal handling. No consideration was given to the low

frequency response of the active low side probes, although this would

enter into an actual calibration. These results are tabulated in table 4.

It is clear that the passive mode would not be usable in the lab, because

of the enormous ratios involved. Very small inductances, which are negligible

at 60 Hz, can cause significant deviations from a pure capacitive behavior

at high frequencies as is shown in figures 45 and 46. These step responses

were calculated for high impedance R1 and R2. The change in step response

due to 1 nH is significant, resulting in an initial transient overshoot of

approximately 60%.

In the laboratory, there are calibration problems as well, but they are more readily addressed than in the field. In principle, it should be

possible to calculate the magnitude of the transient propagating through

the test facility, if the character!' sti c impedances of each section are known. This approach is less reliable than a direct measurement, but should

be investigated more thoroughly, since the combined use of impedance

information and a transient code, such as EMTP, may yield adequate information

with a minimum investment effort.

It is important to carefully define the level of accuracy required for

any given measurement. Based on this level, an error budget for the entire

measurement process can be given and resources allocated appropri ately

.

Based on the work reported here, as well as that of others, there should be

little difficulty in achieving adequate time response to avoid temporal

distortion of electrical signals in the kilohertz to several hundred

megahertz range. However, an adequate means of achieving amplitude

calibration has not been demonstrated.

Methods used to calibrate voltage sensors in large pulse power systems

are described in [19]. Two methods are discussed, both of which requi re access to deenergized equipment. Although useful for comparison, these

methods do not appear to have applicability to the present problem.

61
Figure 45. Step response calculated for plug sensor using circuit in

figure 17 and component values Ci = 1.01 pF, C 2 = 1107 pF,

Rl = 50ft, R

2 = 50ft, and all other components negligible.

62
STEP

RESPONSE

TIME i ns

Figure 46. Step response calculated for plug sensor using

component values listed in figure 45, except Lg = 1 U

H (see fig. 17)

63
Table 4. Calculated ac response for plug sensor

Freq • (Hz) V /V • a

v o'

v m V /V- b

o' in

6.0E + 1 1.904E - 8 1.904E - 8

6.0E + 2 1.904E - 7 1.904E - 7 6.0E + 3 1.9U4E - 6 1.904E - 6

6.0E + 4 1.902E - 5 1.9U2E - 5 6.0E + 5 1.757E - 4 1.757E - 4

6.0E + 6 4.432E - 4 4.426E - 4

6.0E + 7 4.556E - 4 3.840E - 4

6.0E + 8 4.558E - 4 6.808E - 3

a q

= 1.01 pF, C2 = 1107 pF, R| = 50 fi, R 2

= 50 n All other components are negligible (fig. 17).

b

q

= 1.01 pF, C

2 = 1107 pF, R-j, = 50 n, Lg = 1 nH.

All other components are negligible (fig. 17).

Freq. (Hz) V /V • c

o' m V /V- d

v o' in

6.0E + 1 8.865E - 5 8.865E - 5 6.0E + 2 9.113E - 5 9.113E - 5

6.0E + 3 9.115E - 5 9.115E - 5 6.0E + 4 9.115E - 5 9.115E - 5

6.0E + 5 9.115E - 5 9.102E - 5 6.0E + 6 9.115E - 5 9.101E - 5 6.0E + 7 9.115E - 5 7.682E - 6

6.0E + 8 9.115E - 5 1.363E - 3

c C

1 = 1.01 pF, C2

= 1107 pF, Ri = 9 Mq, R

2 = 1 Mn.

All other components are negligible (fig. 17).

d

c

x = 1.01 pF, C

2

= 1107 pF, R

1 = 9 M n, R2 = 1 Mfi, Lg = 1 nH.

All other components are negligible (fig. 17).

64
7. SUMMARY

Three different capacitive probes intended for use in measuring

transient signals in gas-insulated systems were investigated. In a system,

these probes form a capacitor divider but differ from the conventional

capacitor divider in that the high side capacitance does not exist as a discrete component. This complicates the process of characterizing the

probe, since the value of the high side capacitance is determined by the

geometrical relation between the probe and the high voltage electrode.

One probe was a near-field scale model similar to that found in BPA

equipment; a second was based on designs used in pulse power measurements;

while the third was configured for ease of use in existing equipment. The

model sensor was found to have serious shortcomings. Inductance associated

with the assembly resulted in sensor behavior significantly different from

that predicted by a simple equivalent circuit for a capacitive divider.

This behavior was evidenced by large oscillations in the output, and a high

level output voltage. The sensor was found to be useless for transient

measurements

.

A second probe, the plug probe, was similar to those described in the

literature which had been used in both gas and liquid insulated pulse power

systems. This divider had electrical characteri sti cs which agreed, to within

the measurement capabilities employed, with those calculated by using

an ideal model for a capacitive divider. There was no evidence of device

oscillation "below 1 GHz, and no indication of low side inductance.

A third sensor had a disk geometry, and could be installed directly in

place of the existing BPA sensors. This disk sensor compared favorably

with the plug sensor, but could not be evaluated as thoroughly because of

geometrical constraints in the test facility. The design of this sensor

was not optimized and all laboratory tests were made on a "brass board"

system.

A characteristic of capacitive dividers is the RC decay associated

with the step response of the system. For many passive systems, this

restricts their usefulness to waveform durations of less than 1 pS. The

time constant can be increased by orders of magnitude by using active

electronics in the low side. This may be done by employing fast buffer

amplifiers (cable drivers) or commercial FET input probes. When this is

done, there is some loss of rise time and decrease in reliability. For

gas-insulated systems, the rise times achieved may be adequate, if primary

interest is in observing waveforms which extend over the several hundred kHz

to MHz range. If specific information about the rise time of the primary

breakdown process is desired, then a passive probe can be used since long-term

waveforms are not of interest.

The application of discrete component equivalent circuits was useful

in understanding sensor behavior. These models provided upper limits to

various stray inductances associated with the plug sensor. This knowledge

is important in considering how to calibrate the sensor, since the low side

impedance must be thoroughly characteri zed in the laboratory.

65
The problem of divider calibration remains largely unsolved. Calculations

based on discrete component equivalent circuits show very small inductances

in divider low sides can cause significant deviations from low frequency

ratios at near GHz frequencies. Experimental and theoretical work was done using the approach of determining in the laboratory all relevant

information about the divider except Cj_. In principle, the magnitude of

the voltage pulse propagating in the transmission line test facility can be

calculated, but because of uncertainties in reflection coefficients at

impedance mismatches, there is uncertainty in this procedure. A more accurate

calculation would involve use of a wel 1 -characteri zed line, as determined by use of time domain ref lectometry and a transient code such as EMTP. This

approach was not explored in any detail, but does not involve any extension

of existing technology. Since C]_ and C?_ can be accurately determined in

the laboratory, a method exists for comparing experiment and theory.

The results of calculations of the steady-state response of the

capacitive sensors over a wide frequency range demonstrate the futility of

trying to do a calibration at 60 Hz unless inductance in the system can be

kept well below 1 nH. The usefulness of this approach using active low

side electronics was only explored in a preliminary way, and would require

extensive effort for a comprehensive evaluation.

The experimental line as it presently exists is far from optimum. The

facility would be much more useful if the impedance mismatches could be

reduced to provide a more uniform impedance. In the high voltage mode, a cable extension to the line could provide a high voltage square wave with a duration of more than one microsecond without significant droop. The impedance

of the overall line should be determined by using ref lectometry methods to

provide data necessary for computer modelling of pulse propagation on the line.

Since useful sensor designs have been determined, there should be

emphasis placed on developing calibration methods for both laboratory and

field application. This will require experimental and theoretical effort.

The usefulness of active electronics in the low side of the sensor has

not been thoroughly investigated. Although this approach does extend the

low frequency response of the system, the overall system capability has not

been looked at. Information regarding system stability, thermal response

and reliability is lacking.

Although the individual components of a measurement system consisting

of sensor, low side electronics and optical links can be studied separately,

it is advisable to determine the integrated system package response to

signals of the type encountered in the field. The laboratory setting is an ideal setting in which to do this.

8. ACKNOWLEDGMENTS

Prof. Robert Harrington and Mohamed El-Faham of George Washington

University contributed much of the information contained in section 2.1 and

provided useful discussions of transient potentials on transmission line

sheaths

66
9. REFERENCES

[1] Workshop on User Experience With Gas Insulated Substations, EPRI report

EL-2189, Dec. 1981.

[2] Bibliography of Gas Insulated Substations, prepared by the Working Group

on Gas Insulated Substations, paper 32WM169-1, IEEE PES Winter Meeting,

Jan. 31-Feb. 5, 1982.

[3] S. Narimatsu, K. Yamaguchi, S. Nakano, and S. Maruyama, Interrupting

Performance of Capacitive Current by Disconnecting Switch for Gas Insulated

Switchgear, IEEE Trans. PAS 100 , pp. 2726-2732, 1981.

-[4] S. A. Boggs, F. Y. Chu, N. Fukimoto, A. Krenicky, A. Plessl, and D. Schlicht,

Disconnect Switch Induced Transients and Trapped Charge in Gas Insulated

Substations, paper 82WM003-2, IEEE PES Winter Meeting, Jan. 31-Feb. 6, 1982.

[5] L. Blahous and T. Gysel, Mathematical Investigation of the Transient

Overvoltages During Disconnector Switching in GIS, paper 83WM109-6,

IEEE PES Winter Meeting, Jan. 30-Feb. 4, 1983.

[6] N. Fujimoto, E. P. Dick, S. A. Boggs, and G. L. Ford, Transient Ground

Potential Rise in Gas Insulated Substations - Experimental Studies,

paper 82WM011-5, IEEE PES Winter Meeting, Jan. 31-Feb. 5, 1982.

[7] W. Pfeiffer, Ultra-High-Speed Methods of Measurement for the Investigation

of Breakdown Development in Gases, IEEE Trans. Dist. and Meas., IM. 26,

pp. 367-372, 1977.

[8] S. A. Boggs, G. L. Ford, and R. C. Madge, Coupling Devices for the

Detection of Partial Discharges in Gas Insulated Switchgear, IEEE Trans.

PAS-100 , pp. 3969-3973, 1981.

[9] IRR-IMS Group, "Facing UHV Measuring Problems," Electra, No. 35,

pp. 157-254, July 1974.

[10] K. Tokoro, Y. Harumoto, h'. Yamamoto, Y. Yoshida, H. Mukae, Y. Ohno,

M. Shimada, and Y. Ida, Development of Electronic Potential and Current

Transducers Suitable for Gas Insulated Switchgear and Adequate for

Application to Substation Digital Control System, paper 82WM172-5,

IEEE PES Winter Meeting, Jan. 31-Feb. 5, 1982.

[11] D. Pellinen and M. S. D. Capua, Two-Megavolt Divider for Pulsed High

Voltages in Vacuum, Rev. Sci . Inst. 51 , pp. 70-73, 1980.

[12] D. F. McDonald, C. J. Benning, and S. J. Brient, Subnanosecond Rise Time

Multikilovolt Pulse Generator, Rev. Sci. Inst. 36_, pp. 504-506, 1965.

[13] G. E. Leavitt, J, D. Shipman, Jr., and I. M. Vitkovitsky, Ultrafast

High Voltage Probe, Rev. Sci. Inst. 36_, pp. 1371-72, 1965.

[14]

A. J. Schwab and J. H. W. Pagel , Precision Capacitive Voltage Divider

for Impulse Voltage Measurements, IEEE Trans. PAS 91 , pp. 2376-2382, 1972.

67
[15] E. Kuffel and M. Abdullah, High Voltage Engineering, Pergamon Press,

1970.

[16] A. J. Schwab, High-Voltage Measurement Techniques, M.I.T. Press, 1972.

[17] A. Courts, BPA, private communication.

[18] N. W. Harris, High Voltage Probe for Liquid Imersion in Measurement of

Electrical Quantities in Pulse Power Systems, R. H. McKnight and

R. E. Hebner, eds., NBS Special Publication 628, pp. 20-25, 1982.

[19] M. Wilkinson and E. Chu, Calibration of Capacitive Voltage Probes

in Water-Dielectric, High-Power Pulse Generators in Measurement of

Electrical Quantities in Pulse Power Systems, R. H. McKnight and

R. E. Hebner, eds., NBS Special Publication 628, pp. 59-68, 1982.

[20] N. Fijimoto, S. A. Boggs, and R. C. Madge, Measurement of Transient

Potentials in Coaxial Transmission Lines Using Coaxial Dividers j_n

Measurement of Electrical Quantities in Pulse Power Systems,

R. H. McKnight and R. E. Hebner, eds., NBS Special Publication 628,

pp. 69-79, 1982.

[21] E. Nolting, R. Martin, and M. Ruppalt, Electrical Measurement

Techniques Used at the Casino Facility i_n_ Measurement of Electrical

Quantities in Pulse Power Systems, R. H. McKnight and R. E. Hebner, eds.,

NBS Special Publication 628, pp. 118-132, 1982.

[22] G. P. Metzger, M. Riednger, F. M. Schmitt, and G. L. Sutter, Measurement

of Nanosecond Very High Voltage Pulses, Nucl. Inst. Methods, _61_, pp. 226-228, 1968.

[23] W. A. Edson and G. N. Oetzel , Capacitance Voltage Divider for High-Voltage

Pulse Measurement, Rev. Sci. Inst., J52_, pp. 604-oUb, 1981.

[24] C. A. Ekdahl, Voltage and Current Sensors for a High-Density Z-Pinch

Experiment, Rev. Sci. Inst., _51_, pp. 1645-1651, 1980.

[25] T. Haroda, Y. Aoshuma, M. Harada, and K. Huva, Development of HighPerformance Low Voltage Arms for Capacitive Voltage Divider, paper 42.14

presented at the Third International Symposium on High Voltage Engineering,

Milan, August 28-32, 1979.

[26] C. Trivelpiece, R. Richardson, J. Shaumer, and J. B. Smith, Digital

Correction of Cable Attenuation Losses, i_n_ Measurement of Electrical

Quantities in Pulse Power Systems, R. H. McKnight and R. E. Hebner, eds.,

NBS Special Publication 628, pp. 381-391, 1982.

[27] An application of a fast buffer amplifier was described by Dr. W. J. Sargeant

i_n_ Measurment of Electrical Quantities in Pulse Power Systems, R. H. McKnight

and R. E. Hebner, eds., NBS Special Publication 628, pp. 170-173, 1982.

68
NBS-114A (REV. 2-80

U.s. DEPT. OF COMM.

BIBLIOGRAPHIC DATA

SHEET (See instructions

)

4. TITLE AND SUBTITLE

PUBLICATION OR

REPORT NO.

2. Performing Organ. Report No. 3. Publication Date

NBSIR 83-2753 August 1983

EVALUATION OF TRANSIENT MEASUREMENT METHODS

IN GAS-INSULATED TRANSMISSION LINES

5. AUTHOR(S)

Ronald H. McKnight and Howard K. Schoenwetter

6. PERFORMING ORGANIZATION (If joint or other than NBS, see in struction s) 7. Contract/Grant No. national bureau of standards

DEPARTMENT OF COMMERCE 8. Type of Report & Period Covered

WASHINGTON, D.C. 20234

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State, ZIP)

Prepared for

Bonneville Power Administration

P. 0. Box 3621

Portland, Oregon 97208

10.

supplementary notes

Document describes a computer program; SF-185, FlPS Software Summary, is attached.

11.

ABSTRACT (A 200-word or less factual summary of most significant in formation. If document includes a si gn i fi cant bibliography or literature survey, mention it here)

Capacitive sensors suitable for measuring transients in gas-insulated

transmission lines have been studied in the laboratory. Measurements of

the step response of three different sensors were made with a test line

using both low voltage (200 V) and high voltage (10 k V ) signals. Sensor

designs were based on those used in pulse power measurements. The use of

active electronics at the sensor output in the form of fast buffer

amplifiers or commercial FET input probes was investigated as a means of

extending low frequency cutoff. Lumped parameter models were used to

provide theoretical analysis of experimental results.

12.

KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolon s)

capacitor sensors; gas-insulated equipment; high voltage;

transient measurements; transmission lines.

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