NBSIR 73-335

CALIBRATION OF RADIO RECEIVERS

TO MEASORE BROADBAND INTERFERENCE

Ezra B. Larsen

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

September 1973

Final Report, Phase I

Prepared for

Calibration Coordination Group

Army /Navy /Air Force

NBSIR 73-335

CALIBRATION OF RADIO RECEIVERS

TO MEASORE BROADBAND INTERFERENCE

Ezra B. Larsen

Electromagnetics Division

Institute for Basic Standards

National Bureau of Standards

Boulder, Colorado 80302

September 1973

Final Report, Phase I

Prepared for

Calibration Coordination Group

Army/Navy /Air Force

u s. DEPARTMENT OF COMMERCE, Frederick B. Dent. Secretary

NATIONAL BUREAU OF STANDARDS. Richard W Roberts Director
V
CONTENTS

Page

1. BACKGROUND 2 2. INTRODUCTION 3

3. DEFINITIONS OF RECEIVER BANDWIDTH AND IMPULSE

STRENGTH (Or Spectral Intensity) 5 3.1 Random noise bandwidth 5 3.2 Impulse bandwidth in terms of receiver

response transient 7 3.3 Voltage-response bandwidth of receiver 8

3.4 Receiver bandwidths related to voltageresponse bandwidth 9

3.5 Impulse strength in terms of random-noise

bandwidth 10

3.6 Sum-and-dif ference correlation radiometer

technique 10

3.7 Impulse strength in terms of Fourier transforms 11

3.8 Impulse strength in terms of rectangular

DC pulses 12

3.9 Impulse strength and impulse bandwidth in

terms of RF pulses 13

4. EXPERIMENTAL PROCEDURE AND DATA 16

4.1 Measurement of receiver input impedance and

phase linearity 16

4.2 Calibration of receiver as a tuned RF voltmeter 20

4.3 Stepped- frequency measurements of receiver

bandwidth 23

4.4 Calibration of receiver impulse bandwidth

with a baseband pulse generator 37

iii
CONTENTS [Continued)

Page

4.5 Calibration o£ receiver impulse bandwidth

with a pulsed-RF source 38

5. DISCUSSION OF IMPULSE INTERFERENCE AND FOURIER

TRANSFORMS 50

6. SUMMARY AND CONCLUSIONS —-^58

7. ACKNOWLEDGMENTS 6 7

8. REFERENCES 68

iv
LIST OF FIGURES

Page

Fig. 1. Measured Input VSWR of FIM-1 Receiver Tuned to

400 MHz, Signal Frequency = 100 to 1200 MHz 17

Fig. 2. Measured Input VSWR o£ FIM-1 Receiver Tuned to

400 MHz, Signal Frequency = 395 to 407 MHz 18

Fig. 3. Variation of Receiver Input Impedance and Phase

Angle Across the Receiver Passband 19

Fig. 4. Block Diagram of Instrumentation for Calibrating

a Receiver to Measure CW Voltage and SteppedFrequency Bandwidth 21

Fig. 5. Measured Insertion Loss vs Frequency of the

27 dB and 47 dB L-Pads 22

Fig. 6. Stepped-Frequency Measurement of Receiver Bandwidth at 300 MHz, dB Response vs Signal Frequency, FIM-2, Wide Bandwidth 26

Fig. 7. Stepped-Frequency Measurement of Receiver Bandwidth at 300 MHz, Response Voltage vs Signal

Frequency, FIM-2, Wide Bandwidth 27

Fig. 8. Stepped-Frequency Measurement of Receiver Bandwidth at 300 MHz, Squared Response Voltage vs Signal Frequency, FIM-2, Wide Bandwidth 28

Fig. 9. Stepped-Frequency Measurement of Receiver Bandwidth at 500 MHz, FIM-2, Wide Bandwidth 29

Fig. 10. Stepped-Frequency Measurement of Receiver Bandwidth at 700 MHz, FIM-2, Wide Bandwidth 30

Fig. 11. Stepped-Frequency Measurement of Receiver Bandwidth at 900 MHz, FIM-2, Wide Bandwidth 31

Fig. 12. Stepped-Frequency Measurement of Receiver Bandwidth at 300 MHz, dB Response vs Signal Frequency, FIM-2, Narrow Bandwidth 32

Fig. 13. Stepped-Frequency Measurement of Receiver Bandwidth at 300 MHz, FIM-2, Narrow Bandwidth 33

V
LIST OF FIGURES, continued

Page

Fig. 14. Stepped- Frequency Measurement o£ Receiver Bandwidth at 500 MHz, FIM-2, Narrow Bandwidth 34

Fig. 15. Stepped-Frequency Measurement of Receiver Bandwidth at 700 MHz, FIM-2, Narrow Bandwidth 35

Fig. 16. Stepped-Frequency Measurement of Receiver Bandwidth at 900 MHz, FIM-2, Narrow Bandwidth 36

Fig. 17. Impulse Strength vs Frequency of NBS Baseband

Impulse Generator 39

Fig. 18. Block Diagram of Instrumentation for Calibrating

Receiver Impulse Bandwidth with a Standard Baseband Pulser 40

Fig. 19. Check of Receiver Linearity for Impulse Interference at a Tuned Frequency of 400 MHz 41

Fig. 20. Graph of Receiver Linearity for Impulse Interference at 400 MHz 42

Fig. 21. List of Measured Impulse Bandwidths of FIM-2

Receiver, Using a Standard Baseband Impulse

Generator 43

Fig. 22. Block Diagram of Instrumentation for Calibrating

Receiver Impulse Bandwidth with a Pulsed-RF

Source 47

Fig. 23. Waveform of the Pulsed-RF Burst as Measured

with a Sampling Oscilloscope 48

Fig. 24. List of Measured Impulse Bandwidths of FIM-2

Receiver, Using a Pulsed-RF Source with Effective Time Duration of 0.1 y Seconds 49

Fig. 25. Summary of Measured Bandwidths of the FIM-2

Receiver 60

Fig. 26. Graphs of Measured Bandwidths of the FIM-2

Receiver 61

Fig. 27. Table of Ratios of Measured Bandwidths of the

FIM-2 Receiver 66

vi
CALIBRATION OF RADIO RECEIVERS

TO MEASURE BROADBAND INTERFERENCE

ABSTRACT

This report covers one phase of a project to improve the accuracy for calibrating field-strength meters

to measure broadband signals, especially impulse interference. The amplitude of a narrowband signal is commonly expressed in pV, and a broadband signal is expressed in yV/MHz bandwidth. A future report C2nd

phase) will deal with standards and techniques for

producing a known impulse field to calibrate an

antenna-receiver system in yV/(m*MHz). The technique

used for initial calibration of a receiver as an RF

voltmeter is new, employing a special L-pad with

48 9. input resistor and 2 Q. output resistor across

the receiver. The input power across 50 Q, is leveled,

producing a low -impedance , cons tant -voltage source

at the receiver input terminal.

Broadband interference is generally divided into

two types: (a) random noise is best characterized by its

power spectral density, and (b) impulse interference

(uniform periodic pulses) is best quantified by its impulse strength. Several possible approaches were

evaluated for defining and measuring various bandwidths

of a receiver. The response of a receiver to impulses,

as measured with a peak-reading detector, is proportional

to the product of receiver impulse bandwidth and signal

impulse strength. Of the two types of impulse sources

evaluated here (pulsed DC and pulsed RF) , the latter

appears to be the most accurate and repeatable. We were

able to produce a clean RF pulse, choosing an effective

duration of 0.1 ys in our experiments, which covered a frequency range of 100 to 1000 MHz. The worst-case

uncertainty of this standard source for calibrating

receiver impulse bandwidth is ± 0.7 dB

.

Key words: Broadband interference; field strength

meter; impulse standards; receiver bandwidth calibration; spectral intensity.
1. BACKGROUND

This report covers one phase o£ a project at the National

Bureau o£ Standards (NBS) to improve the overall accuracy for

calibrating field strength meters, concentrating on accurate

measurement of broadb and interference. The present report

deals with measuring conducted radio - frequency (RF) signals,

that is, signals at the antenna input terminal of a measuring

receiver. The amplitude of a narrowband signal is commonly

expressed in units of microvolts (yV) , whereas a broadband

signal is expressed in microvolts per megahertz bandwidth

(yV/MHz) . A subsequent report (2nd phase) will deal with

development of standards and accurate techniques for calibrating a radio interference meter or field intensity meter

(RI/FI meter or FIM) as a system ; that is, calibration of

the antenna, receiver and associated hardware while connected

in a normal-use configuration. The impulse response of a measurement system is affected by impedance mismatches, such

as between the antenna and receiver, or between a calibrating

impulse generator and receiver. Previous workers in the area

of broadband interference have generally ignored these system

effects on FIM calibration.

The second phase of this NBS project (future report) involves the design of a pulsed RF soutce and broadband antenna

for radiating a standard impulse field of known amplitude and

spectral characteristics. The amplitude of an electric field

is commonly expressed in units of microvolts per meter for

a narrowband signal and microvolts per meter per MHz bandwidth (yV/(m»MHz) for a broadband signal. The objective of

this second phase is to make meaningful calibrations of field

strength meters for measuring radiated impulse - interference

fields. This is analogous to cal ibrating the "antenna coefficient" or field strength "antenna factor" for measuring

sinusoidal field strength. That is, a calibration factor

is determined experimentally at a field site for each frequency at which the field strength meter will be used. For

frequencies up to 1000 MHz this tan be done by comparing the

measured antenna/ receiver system response to that of a standard open-circuit dipole placed at the same position as the

FIM dipole. The field strength factor thus applies to the

measuring system as connected in its notmal-use configuration.

It would be desirable to also use this system approach for

calibrating a receiver to measure impulse- interference fields.

It would then not be necessary to consider separately the

errors of measured receiver impulse bandwidth, errors caused

by receiver and antenna phase shift across the passband, or

errors due to impedance mismatches at various points in the

system.

2
When measuring broadband interference it is not possible

to use the same "antenna factor" used for sinusoidal fields,

which has a value that is independent of receiver bandwidth.

Receiver response to impulse interference, however, is proportional to the receiver bandwidth. Proper calibration of

the effective impulse bandwidth of a receiver thus makes

possible an absolute measurement of conducted impulse interference. This receiver with calibrated bandwidth can also

be used, in turn, to measure the impulse strength or spectral

intensity of a working impulse generator, such as the internal

impulse- generator source of an FIM. However, the terms

"impulse strength" and "spectral intensity" are variously

defined, and the measured value often depends on the instrument

used to measure it. Further, the "impulse bandwidth" of a receiver depends in important ways on factors other than the

usual voltage -response -vs .- frequency type of bandwidth. These

topics have been investigated at NBS and are the subject of

this report.

2. INTRODUCTION

In this report the term "receiver" is used to designate

a frequency -select ive voltmeter used to measure RF signals.

Generally the receiver is basically a superheterodyne communications-type receiver with one or two additional values of

time constant in the detector. Three types of detectors are

commonly found on an interference meter, namely average, peak

and quasi-peak detectors. The average -of -envelope detector

indicates a time-average voltage of the detected output from

the IF amplifier. The peak detector indicates the maximum

voltage value (long time constant) of the IF amplifier output.

At present no commercially-available FIM employs a power or

RMS detector. No discussion is given in this report of

amplitude probability distribution (APD) or other schemes

used to quantify mixtures of the various types of interference.

The receiver is one part of the overall measurement system

(designated field strength meter, interference meter, FIM or

RI/FI meter) which also contains antennas, step attenuators,

calibrating generators, various metering circuits and output

indicators

An RF signal is considered to be narrowband or broadband

depending on whether its frequency spectrum is less than or

greater than the bandwidth of the receiver. The overall bandwidth of an FIM is determined essentially by the selectivity

of the IF amplifier strip and is typically of the order of

0.1% of the receiver tuned frequency. The term "CW" stands

3
for a s ingle - frequency sinusoid of constant amplitude. The

term "impulse" is used for a baseband (DC) pulse or an RF

burst (pulsed RF) whose time duration is short compared with

the period of the receiver circuit which responds to the impulse. This period is generally approximately the reciprocal

of the IF bandwidth.

A common practice is to divide broadband interference

into two categories, which may be defined as follows: (1) Random noise consists of a large number of overlapping signals

with random amplitude, frequency and relative phases. Examples

are thermal noise and shot noise. (2) Impulse noise is due to

a single short pulse of voltage, or repetitive pulses, causing

a signal with systematic relative phase and amplitude vs. frequency. Examples are switching transients, automobile ignition noise and interference from radar transmitters. The

response of a receiver to either type of broadband signal is

determined (partially) by the receiver bandwidth. This is

because all frequency components of the input signal which

fall outside the receiver bandwidth are rejected, or at least

amplified less than the frequency components at the center

of the passband.

Differences between the possible types of RF signals can

be illustrated by comparing receiver response (indicated level)

as a function of receiver bandwidth. For CW the response

amplitudes for a peak-voltage, average -volt age , or RMS-voltage

detector is independent of the receiver bandwidth. For random

noise the average -voltage and RMS response values of an FIM

are proportional to the square root of bandwidth. For im^

pulse noise of low repetition rate, the average voltage is

typically independent of bandwidth but the peak value is

proportional to bandwidth (among other things) . This proportionality factor makes it convenient to define an impulse

bandwidth for the receiver. Actually, the response of a tuned receiver to an impulse, as measured with a peak-reading

detector, depends not only on bandwidth but also on the shape

of the bandpass curve and phase cliaracterist ics of the receiver. The response of ideal interference meters to various

inputs, as a function of bandwidth, is summarized in the following table.

4
Type of

Detector

Type of RF Signal

Sine wave Random noise

rer iodic

impulses

Average

( volt ag

Receiver

cation

indi - is : Independent of

bandwidth (BW)

Proportional

to /B¥

Independent

of BW

Peak

(volt ag

Receiver

cat ion

indi - is : Independent of

bandwidth (BW)

Proportional

to BW

Power Receiver

cation

indi - is : Independent of

bandwidth (BW)

Proportional

to BW

Proportional

to BW

RMS

(vol tag;e)

Receiver

cation

indi - is : Independent of

bandwidth (BW)

Proportional

to y/M

Proportional

to /BW

In most common approaches to impulse measurement, the

impulse bandwidth of the receiver is a calibration factor

determined by adjusting the output of a CW generator to give

the same indication on the receiver output meter as the peak

response to an impulse generator of known characteristics.

The impulse bandwidth in MHz is then the output of the CW

generator in yV, divided by the output of the impulse generator in yV/MHz . Some commonly-used definitions relating to

broadband interference are discussed in the next section.

3. DEFINITIONS OF RECEIVER BANDWIDTH AND IMPULSE STRENGTH

(OR SPECTRAL INTENSITY)

The "effective" or "equivalent rectangular" passband of

a given receiver is different for random and impulse noise.

Also, the numerical value of impulse bandwidth depends on

the definition chosen for this quantity. A summary of various

definitions for bandwidth and the corresponding equations for

impulse strength are given in this section. We do not necessarily endorse all of these definitions; in fact we consider

the use of some of them to be ill-advised. The receiver gain

is assumed to have been calibrated in terms of the root -meansquare (RMS) voltage value of an applied sine wave.

3.1 Random noise bandwidth or squared-volt age -response bandwidth or "power" bandwidth of receiver [1] and [2]

.

5
[V(f)]

f = 0

1 0

EQUAL AREA, A

(a) Equivalent rectangular bandwidth of receiver, as determined from a point-by -point measurement and

analys is

.

where

(BW)

ran

/ [V(f)]^ df

0

[V^(f)]^

Area under squared- volt age -response curve

Square of maximum voltage response

(1)

(b) Power spectral density or random-noise spectral intensity, as measured by a receiver whose bandwidth

has been calibrated by eq. (1) above.

(S)

[VJ£)]

ran (G 2) (BW) (R ) ^ o ^ ^ ^ ran ^ o-'

watts

hertz

(2)

G = CW gain of receiver (maximum) at tuned frequency

= Receiver input resistance (50 f2) . A discussion of these expressions is given in section

4.3 of this report. The random-noise bandwidth of a receiver

should technically be defined in terms of the power response

for a Gaussian signal which is flat with frequency across the

receiver passband. The power delivered to a receiver within

its passband is given by

P = (S) (BW) ^ ran ^ ^ ran (3)

The definition generally used for receiver random-noise

bandwidth is given in terms of a fictitious, rectangular

6
passband which has the same area as the actual receiver

passband. As indicated by the sketch on page 6, this bandwidth is given by the expression

(BW)

ran

(£2-f^) = Area A/[V^(f)]

As described in section 4 o£ this report, the area A is

determined at NBS with high precision by using a steppedfrequency approach and an automatic network analyzer.

3.2 Impulse bandwidth in terms of receiver response transient

LlJ, [2J, 13J, L4J, [5J and MIL-STC)-463.

v( t)

ENVELOPE = V(t) V (t)

m EQUAL AREA

/(BW)-

1 m p

(a) Equivalent rectangular bandwidth o£ receiver, as determined from the envelope of the receiver impulse

response transient.

(b)

(BW)

V (t) m^ ^ imp 00/ V(t)

o

(4)

dt

Maximum voltage of receiver response envelope

Area under response envelope curve

Strength of impulse signal applied to the antenna

input terminal , as measured by a receiver whose

impulse bandwidth has been calibrated by eq. (4).

V (t) m^ ^ imp

(G ).rBW) . ^ o^ " imp

volts

hertz

or volt 'Seconds

.

(5)

That is, impulse strength of an ideal impulse is defined

as the peak response of an ideal amplifier to this pulse

divided by the bandwidth. Also,

7
O

This approach for measuring impulse bandwidth is usually

done by photographing an oscilloscope display of the receiver

IF response to identical repetitive pulses. The positive

area under the envelope curve of the receiver response

transient can be measured by counting squares or with a planimeter or it can be integrated electronically. In any

case, the measurement precision is somewhat limited. There

are also fundamental errors with this definition as explained

in [6] and the discussion of section 5. According to Palladino

[7] "The preceding method is used simply because no other accurate economic convenient method is currently available."

It can be shown mathematically that the impulse strength of

an instantaneous DC pulse (delta function) in volts per hertz

is numerically equal to twice the area of the pulse in

volts • seconds . This means that the area A under the receiver

response curve [for the ideal case and after normalizing for

receiver gain) would be double the area of the original

applied pulse, in volt • seconds

.

_ Area A

G

0

3.3 Voltage -response bandwidth or swept -CW bandwidth of

receiver [1] , [T] and [5] . This is sometimes given

( incorrectly) as an alternative definition of impulse

bandwidth.

(a) Equivalent rectangular bandwidth of receiver, as determined from the CW volt age - response curve

(swept signal frequency or swept bandpass filter)

.

/ V(f) df

^^^)voltage ^

'

^ CBW).^p C6)

m*- ^

_ Area under voltage - response curve , vol ts »hert

z

Maximum detected voltage at frequency £

8
Cb) Strength of impulse signal applied to the antenna

input terminal, as measured by a receiver whose

impulse bandwidth was calibrated by eq. (6).

(G ) (BW)

.

^ o ^ ^ imp

The value o£ bandwidth when using this definition can

readily be measured with a s tepped- frequency CW approach,

similar to that of random noise bandwidth. The numerical

integration is more accurate for the random noise or power

(voltage squared) case of section 3.1 than for the voltage

case of section 3.3. This is due to the inherent error in

integrating the area under long tails. In either case the

numerical integration can be programmed on an automatic network analyzer. The peak value of receiver response transient

V^(t) can be measured with a peak reading voltmeter, such as those found on military and commercial interference meters.

There is a fundamental error if eq . (6) is used to define impulse bandwidth, as explained on page 7 of [1] . This

definition would be correct only if the phase is linear across

the receiver passband. Since phase shift is present in a practical amplifier (see section 4.1 of this report for graphs

of data .taken for a typical FIM .receiver) the impulse bandwidth should be defined differently. According to Geselowitz,

page 7 of [1], "since ^ is, in general, not linear over the

passband, the integrated bandwidth cannot be used as the

effective impulse bandwidth. For example, in the case of a single-tuned resonant circuit, the integrated bandwidth becomes infinite! ... The integrated bandwidth (of eq. (6)) is

an upper limit which may be approached in some cases, e.g.,

as the number of tuned stages increases." A similar comment

can also be found on page 72 of [8]

.

3 . 4 Receiver bandwidths related to the voltage -response

bandwidth described in section 3.3.

3 dB Bandwidth = Frequency width of the receiver selectivity

curve at the points where the response voltage is 3 dB below the maximum response.

The 6 dB , 10 dB , and 20 dB bandwidths are defined in a similar

manner.

9
3 . 5 Definition of impulse strength in terms of random-noise

bandwidth and stepped CW measurements.

This approach was proposed by Andrews [9] and has been

used for a prototype measurement system. The basic equation

for the so-called "root -mean-square spectral intensity" is

(S) . ^ ^ imp

1 / (^^o^ ^^0^

. (8)

G. 1/ (PRF) (BW)^^^

where: (S)^j^p = Strength of impulse generator being calibrated

P = Average power output of calibrating system

° into R

o

= System load resistance (50 $7 power meter)

= System tuned- frequency CW gain

PRF = Pulse repetition frequency of impulse generator

(BW) = Random noise bandwidth, as defined by eq. (1)

,

ran , page 6.

It can be seen that this approach defines impulse strength

in terms of random noise bandwidth (obtained ftotn CW measurements) and measurements of CW power and CW voltage. The system described by Andrews measures the bandwidth on a steppedfrequency basis, establishing the RF voltage at each frequency

by means of a low-resistance RF micropot. An impulse generator calibrated with eq. (8) can then be used to calibrate the

impulse bandwidth of an unknown receiver by the usual expression of eqs. (5) and (7):

V (t)

^"^P (G ) (S) - ^ o^ ^ ^ imp

3 . 6 Sum- and-diff erence correlation radiometer technique

[10] and [11].

This technique was developed at NBS to calibrate the power

spectral density of random noise in watts/Hz, by radiometric

means. The approach can be used to measure a random noise

generator and, in turn, calibrate the random noise bandwidth

of a receiver. Also, the strength of an unknown repetitive

impulse generator can be compared against a thermal white-noise

10
source whose power spectral density is known, making use of

the pulse repetition rate and impedance o£ the impulse source

being calibrated. It can be seen from the table on page 5 that

the RMS response of a receiver is proportional to /BW for either

random or impulse noise.

However, this approach suffers from the required complexity of instrumentation and the difficulty of measuring

the weak power output of an impulse generator. It should be

mentioned that most of the "internal" impulse generators of

FIM's have low repetition rates, typically about 60 Hz. The

average power output of an impulse generator is thus extremely

low since the duty cycle may be less than 0.00001%. According

to Reeve, page 28 of [11], "it would appear that radiometric

techniques, while having potentially high accuracy, are not

desirable for general measurements of impulses. Radiometers

measure total power and respond to artifacts between the

major impulses which means they will be measuring something

different than the typical impulse receiver which responds

only to the peak amplitude of the largest signal appearing

at its detector. ... A radiometric technique would be of

great value in cross-checking the accuracy of another type

of impulse calibration system providing the impulse signal

source used as a transfer standard is known to be free of

artifacts and the repetition rate is kept high to maintain

an adequate signal to noise ratio."

3 . 7 Definition o f impulse strength in terms of Fourier

trans forms [6 ] , [ 12 ] and [13]

.

Another practice is to define impulse strength in terms

of the absolute value of the Fourier transform of a single

time-domain pulse. See discussion in section 5 of this report. In this method, repetitive pulses from a baseband

impulse generator are measured (digitized) in the time domain,

for example with a sampling oscilloscope, and the Fast

Fourier Transform is calculated with a digital computer. The

impulse generator with "known" impulse strength (S)

^^^p

is

then applied to a receiver to calibrate its impulse bandwidth.

The mathematical expression used is the basic definition

of eq. (9), as follows:

V (t)

^"^P (G ) (S) . ^ o imp

11
A known 50 Q. impulse generator using a step-recovery

diode source was fabricated and tested at NBS [12] . Experimental measurements made with this standard impulse generator

are described in section 4.4 o£ this report.

3 . 8 Definition of impulse strength in terms of rectangular

DC pulses [3] and [6]

.

The time and frequency domain properties of repetitive

baseband pulses are indicated in the following sketch. The

vertical lines in the frequency-domain sketch represent the

spectrum V(f) of the time-domain pulses v(t) . That is, the

Fourier components of the impulse signal shown consist of

essentially discrete frequencies. For DC pulses these lines

occur at f=0 , 1/T, 2/T, etc.

TIME DOMAIN

FREQUENCY t)OMAIN

v(t)

PRF

T

An often-made statement is that impulse strength is equal

to double the area under the time domain curve of a baseband

pulse. This is strictly true only for an idealized deltafunction impulse, in which the pulse duration is zero and

the pulse height is infinite. In this case the frequency

spectrum V(f) has constant amplitude, independent of frequency. For pulses of finite duration the impulse strength

is also often defined as twice the pulse area, in volt'seconds

or volts/hertz. Actually, the frequency spectrum of a single

rectangular pulse is given by

sin TTfx' _ V(f) = (V^T)

^ Trf T

E VqT sine (Trfx) , (11)

12
where: = Pulse amplitude in volts

T = Pulse duration in seconds

£ = Frequency in hertz

sine Trfx = (sin Trfi) / (irfT) .

For short pulses and low frequencies the sine function

approaches 1, and the impulse strength is approximately 2V^t , or twice the pulse area. That is, for low frequencies where

f << 1/T,

(S).j^p " 2V^T, volts/hertz. (12)

Some impulse -generator calibrators are based on this

approximation. It is then essential that the receiver bandwidth be narrow compared with l/x. However, for determining

the strength S of a standard impulse generator it is also

necessary to use pulses which are long enough so that V^t or

pulse area can be measured accurately. As far as the response

of a filter circuit to a pulse is concerned, the actual pulse

is often considered as instantaneous if the time duration of

the pulse is much less than (e.g., < 0.2) the reciprocal of

the frequency of the filter circuit. For an interference

meter this filter circuit is essentially the IF strip of the

receiver.

The impulse bandwidth of a receiver to be used for measuring impulse interference can then be calibrated with this

standard impulse generator in terms of the basic equation

V (t) . V (t) , Wi^^ = .-nL_ - (13)

^"^P (G ) (S) . 2G V^T

o'^ ^ imp o 0

3 . 9 Definitions of impulse strength and impulse bandwidth in

terms of RF puls es [14], [15] and [16].

(a) Impulse bandwidth of receiver, as determined by

applying a pulsed RF with known effective time

duration t.

^ ^ imp o 1 c

13
where: V (t)

(S)

G = o

imp

^1

=

V_ =

Maximum value of receiver response transient,

as a function of time

CW gain of receiver (maximum) at tuned frequency f^

Calculated impulse strength of calibrating

signal = V-j^T in this case

RMS voltage of CW signal (during the RF pulse)

applied to the receiver antenna-input terminal

RMS voltage of CW signal (during the RF pulse)

at the receiver detector, after amplification

in the receiver

Time duration of rectangular RF burst; if

the pulse is not rectangular, t is the "equivalent rectangular" time width.

(b) Strength of impulse signal applied to the

antenna input terminal, as measured by a receiver

whose impulse bandwidth has been calibrated by

eq. (14).

(S)

V (t) m^ ^ imp

G (BW)

.

o ^ ^ imp

(15)

The time and frequency domain properties of repetitive

RF pulses are indicated in the following sketch. Note that

the measuring receiver must be tuned at or near the CW calibration frequency f^ . Also, the above equations assume that

there is no impedance mismatch between the impulse generator

output and the receiver input. These definitions are used

for two different commercial impulse calibration systems,

and apply to MIL-STD 826A. In all these cases the calculable

broadband signal used for calibrating the impulse bandwidth

of an FIM is produced by pulse-modulating a CW source. That

is , near the carrier frequency f^

,

(S) imp volt s/hert z

14
TIME DOMAIN FREQUENCY DOMAIN

The Fourier components of the pulsed RF signal shown consist of discrete frequencies. These spectrum lines occur at

the carrier frequency f ^ , f^ + PRF , f^ - PRF , + 2 PRF

,

etc. The pulse width, t, determines the distance between the

zeroes of the frequency spectrum.

Since the bandwidth of an FIM receiver is normally much

greater than the PRF, it cannot resolve the individual

Fourier frequency -domain components, because several lines are

within its bandwidth. However, the envelope can be resolved,

and it is the peak value of this envelope which is assumed

proportional to impulse strength. This is the peak envelope

value measured by an FIM when switched to "peak" mode. The

requirements for accurate measurement of (BW) . by this

technique are listed as follows: ^

(a) Receiver (BW) ^ > (1.7) (PRF).

This insures that a spectrum analyzer or FIM receiver can

respond to each impulse independently. In other words, the

receiver peak indication will be independent of PRF. According

to page 11 of [15], the value of 1.7 allows the RCVR response

transient to decay to 1% of the peak value between pulses.

(b) Receiver (BW) ^ < (0 . 1 ) /

(

t)

^ ^

^

.

This insures that the receiver bandwidth is small (< 5%)

compared with the main lobe width from (f^-l/x) to (f^+l/i) of

the s inc - function frequency spectrum, thus indicating a true

peak value

.

15
(c) Impulse duty cycle = x/T < 0.05.

This insures that the main lobe width of the frequency

spectrum is broad enough to represent a true broadband signal,

whose amplitude is essentially flat across the passband of

the receiver, [15].

(d) Number of cycles in RF burst > 10.

This insures that the receiver indication (for a peak detector) is essentially the same for incoherent pulses as for

coherent pulses, assuming that the two types of pulses have

equal amplitude and time duration. A coherent burst is defined

as one having an integral number of RF cycles, and gated on at

a zero crossing of the RF sine wave, [11].

. ' - 4. EXPERIMENTAL PROCEDURE AND DATA

Measurements of receiver input impedance and complex reflection coefficient were made for a typical interference

meter, to get an indication of the receiver bandwidth and phase

linearity. A comparison was made of various methods used for

the initial calibration of a receiver as a CW frequency- selective voltmeter. Measurements of receiver bandwidth according

to several different definitions were then performed for several

values of input signal level and receiver attenuator step.

Data for calculating various voltage -response bandwidths and

random-noise bandwidth were taken by a stepped-frequency approach

using an automatic network analyzer. Measurements of impulse

bandwidth were made of two commercial receivers, designated

here as FIM-1 and FIM-2, by two different techniques (pulsed

DC and pulsed RF) . A comparison was made of several approaches

to measuring impulse bandwidth of a receiver, to determine

if a constant ratio exists between the various bandwidths

.

The technique using a calculable pulsed-RF generator as the

calibrating source appears to be the most accurate and repeatable of those we tested. The experimental results are

summarized here.

4 . 1 Measurement of receiver input impedance and phase

1 inearity . An indication of receiver bandwidth characteristics and

input VSWR variation across the passband of a typical field

strength meter is given in figures 1 and 2. Typically, the

receiver bandwidth is much narrower than the measured antenna

bandwidths; this will be discussed in a later report. Steppedfrequency measurements were then made to determine the variation

of receiver input impedance across the passband. A summary of

16
10

9

8

7

6

5

2 -

II—\—[—{—

r

RECEIVER TUNED FREQUENCY

J L J I \ \ L J L

200 400 600 800 1 000

SIGNAL FREQUENCY, MHz

1200

Fig. 1. Measured Input VSWR of FIM-1 Receiver Tuned to

400 MHz, Signal Frequency = 100 to 1200 MHz.

17
396 398 400 402 404

SIGNAL FREQUENCY, MHz

406

Fig Measured Input VSWR o£ FIM-1 Receiver Tuned to

400 MHz, Signal Frequency = 39 5 to 407 MHz.

18
SIGNAL PHASE SIGNAL PHASE

FREQ. R in X in INPUT .•\NGLE , FREQ . R in X in INPUT ANGLE,

OHMS OHMS VSWR DEC MHz OHMS OHMS VSWR n£G

.

399.0 59 .4 -75 4 3 . 67 -51 8 599.0 119 . 1 + 57 . 9 3 .04 2 5.9

. 1 59 . 9 -68 8 3 19 -49 0 . 1 123 . 2 + 46 . 1 2 .87 20.5

. 2 60 .8 -63 0 3 . 01 -46 0 . 2 125 . 0 + 33. 5 2 71 15.0

. 3 61 .8 -57 . 0 2 .73 -42 7 . 3 123 .4 + 19 5 2 .54 9 . 0

. 4 63 . 3 -51 8 2 .50 -39 3 . 4 119 . 7 + 7. 8 2 .41 3 . 7

. 5 65 . 1 -46 5 2 .30 -35 5 . 5 113 . 9 - 2 8 2 . 28 -1 .

4

. 6 66 .9 -42 0 2 15 -32 1 . 6 106 .6 -11 9 2 . 17 -6.3

. 7 69 . 0 -38. 1 2 .03 -28 . 9 . 7 98 . 6 -18. 7 2 .07 -10.7

. 8 70 .5 -34. 4 1 .94 -26 0 . 8 89 .9 -24 . 4 1 .99 -15.2

. 9 71 .5 -30. 7 1 85 -23. 3 . 9 81 . 0 -28. 7 1 92 -19.5

CJDO .ID 72 . 6 -27. 0 1 78 -20 4 c5i)0 .ID 72 .5 -31 . 0 1 87 -23.1

. 1 73 .2 -22 . 7 1 70 -17 . 2 . 1 64 . 2 -31. 7 1 83 - 26 . 3

. 2 74 . 6 -18. 5 1 .65 -13. 9 . 2 57 . 0 -31 . 0 1 80. -28.5

. 3 75 .8 -14. 3 1 61 -10 7 . 3 50 .4 -29. 3 1 78 -30.2

. 4 77 . 3 - 9. 7 1 51 •7 1 . 4 44 .9 -27. 4 1 79 -31.4

.5 78 . 6 - 5 2 1 58 - 3. 8 . 5 40 . 2 -25. 3 1 82 -32.2

. 6 80 .2 - 0. 3 1 60 - 0. 2 . 6 36 . 0 -23 . 1 1 87 -32.7

82 . 0 + 4. 9 1 65 3. 4 . 7 32 . 6 -20 . 9 1 94 -32.6

. 8 84 . 2 +10 . 4 1 72 7 . 1 . 8 29 .4 -18 . 6 2 03 -32.3

. 9 86 _ 7 + 16 . 7 1 83 10 . 9 . 9 26 . 9 -16. 4 2 13 -31.5

401.0 90 .6 + 23. 2 1 98 14 . 4 601.0 24 .4 -14 . 4 2 26 -30.5

799.0 30 .5 + 11. 8 1 78 21 . 1 999.0 28 .3 -18 . 8 2 11 -33.6

. 1 31 .9 + 11

.

0 1 69 19 1 . 1

' 29 .5 -18 . 4 2 02 -31.9

. 2 33 .3 + 10. 2 1 61 17. 1 . 2 31 .1 -17. 8 1 91 -29.8

. 3 34 .4 + 9. 2 1 54 15 . 0 . 3 32 .& -17 . 6 1 83 -28.3

. 4 35 .5 + 8. 1 1 48 12 . 9 . 4 34 .1 -17. 4 1 75 -27.0

. 5 36 .3 + 7. 0 1 43 10 . 9 . 5 35 .6 -17 . 2 1 69 -25.7

. 6 37 0 + 5. 8 1 39 8 . 9 . 6 37 . 5 -17 . 2 1 63 -24.7

. 7 37 .7 + 4. 5 1 35 6 . 8 . 7 39 .0 -17 . 5 1 59 -24.2

. 8 38 .2 + 3. 3 1- 32 4. 9 . 8 40 . 7 -17. 9 1 56 -23 .8

. 9 38 7 + 2. 1 1 30 3. 1 . 9 42 .3 -18 . 6 1 54 -23.7

cao 0 .it> 39 . 3 + 0. 8 1 27 1. 2 44 . 0 -19. 2 1 55 -23.6

. 1 39 .7 - 0. 6 1 26 -0 . 9 . 1 45 .6 -20. 2 1 54 -23 . 9

. 2 40 .0 - 2. 1 1 26 -3. 0 . 2 47 .1 - 21. 2 1 55 -24.3

. 3 40 1 - 3. 8 1 27 -5. 4 . 3 48 , 4 -22 . 6 1 58 -25.0

. 4 40 .0 - S. 3 1 29 -7 . 6 . 4 49 .8 -24. 2 1 62 -25.9

.5 39 .5 - 6. 9 1 33 -9. 8 . 5 51 .1 -26. 0 1 66 -27.0

. D 39 .0 - 8. 2 1 36 -11. 9 , 0 51 . 9 -27. 8 1 72

. 7 38 4 - 9. 5 1 41 -13. 9 . 7 52 .4 -29. 8 1 78 -29.6

.8 37 6 -10. 6 1 45 -15 . 7 .8 52 .6 -31

.

5 1 85 -31 .2

. 9 36 .8 -11. 7 1 50 -17. 6 . 9 52 .8 -34. 0 1 92 -32.8

801 . 0 36 1 -12 . 6 1 55 -19 . 2 1001 . 0 52 .7 -36. 1 2 00 -34.4

Fig. 3. Variation of Receiver Input Impedance and Phase Angle

Across the Receiver Passband

(Note: Circled values = receiver tuiied frequencies)

19
this data is given in the table of figure 3. The signal

level used for these measurements was kept low so that the

receiver would not have an attenuator pad inserted internally

in the RF signal path.

It can be seen from figures 1 to 3 that the receiver

presents a badly mismatched load to a 50 source, such as an

antenna or a calibrating impulse generator, for all frequencies

except those very close to the receiver tuned frequency. Even

the tuned- frequency impedance values are considerably different from the nominal value of 50 + jO ohms. For example,

for maximum receiver indication at 400 MHz the input impedance

is 73-j27 ohms, while zero phase angle occurs at 400.6 MHz,

and a minimum VSWR of 1.5 occurs at 400.4 MHz.

4 . 2 Calibration of receiver as a tuned RF voltmeter . A block diagram of the instrumentation used for all the

stepped- frequency measurements of receiver bandwidth is given

in figure 4. This setup was also used for the required initial

calibration of the receiver as an RF voltmeter. Note that this

is a new technique which does not require prior calibration

of the receiver with an RF Micropot. Instead, it employs a special L-pad with approximately 50 H input resistance and a low output resistance. This pad consists of a 48 f2 series

resistor and a 2.35 Q annular-ring resistor in shunt with the

output. The L-pad is housed in a metal box 1 inch long with a type N coax connector at each end.

The low- impedance source for the receiver (2.35 ^ resistor

in figure 4) supplies a known RF voltage to the receiver

input terminal which is essentially independent of variations

in receiver input impedance. For example, assume a receiver

with a nominal 50 input but with an actual input VSWR of

2:1. The maximum possible variation of receiver input voltage

under these conditions is 0.37 dB or 41. As a comparison, the

receiver input voltage varies up to 3.52 dB or 501 when using

a 50 generator as the receiver source. When using the 27 dB

L-pad, the signal level is adjusted to obtain a power meter

reading (accuracy ±11) of 1 mW to produce a 10 mV signal at

the receiver. The 47 dB L-pad has an output of 1 mV for a power input of 1 mW, as indicated in figure 4.

A graph showing the measured insertion loss of the special

L-pads over the frequency range of 100 to 1000 MHz is given

in figure 5. These readings are average values obtained from

a mult iple -scan approach. As can be seen, the attenuation of

the 27 dB pad varies from 26.6 dB at a frequency of 890 MHz,

to 27.4 dB at 190 MHz, or ± 0.4 dB over the entire decade

frequency range.

20
RF SOURCE.

SIGNAL

GENERATOR OR

AUTOMATIC

NETWORK

ANALYZER

(A. N. A. )

COMPUTER

AND MEMORY

OF A. N , A.

LOW-PASS STEP ATTENUATOR.

0 TO 70 dB,

0.1 dB STEPS

FILTERS

20 dB T-PAD 27 dB L-PAD

m 48 a 1

2.^35

50 POWER METER.

10 yW = 100 mV at RCVR,

1 mW = 1 mV AT RCVR,

0.1 W = 10 mV AT RCVR

|POWER METER SETTINGS IF

I 27 dB L-PAD USED ALONE.

I 10 yW = 1 mV AT RCVR,

I

1 mW = 10 mV AT RCVR,

I 0.1 W = 0.1 V AT RCVR

I I

DIGITAL

VOLTMETER

DC VOLTAGE PROPORTIONAL

TO RECEIVER INDICATION

RECEIVER

TO BE

CALIBRATED

Fig. 4. Instrumentation for Calibrating a Receiver to

Measure CW Voltage and Stepped -Frequency Bandwidth.

21

It should be mentioned that improvements could be made in

the construction and attenuation-vs -frequency characteristics

for this type of attenuator pad. The value of 27 dB for insertion loss was chosen for two reasons : (a) the output impedance

is low compared with a SO 9. receiver, and (b) the conversion

from power meter indication to receiver voltage is simple and

convenient

.

If a power meter is used for leveling the power vs. frequency at the input to the L-pad in figure 4, then the output

voltage across the 2.35 9. resistor is also leveled to within

the limits of figure 5. That is, this pad can be used as a lowimpedance voltage source, giving nearly constant amplitude vs.

signal frequency, for swept -frequency calibration of FIM's.

4 . 3 Stepped- frequency measurements of receiver bandwidth . Graphs of some of the data taken for measurement of

receiver bandwidth are given in figures 6 to 16. All of

these graphs are for the FIM-2 "Interference Analyzer."

This interference meter covers 14 kHz to 1 GHz in 15 tuning

bands. The receiver has a 2-position selectivity switch on

the front panel. The "Wide" bandwidth position has a nominal

3-dB bandwidth of 500 kHz and the "Narrow" position a nominal

bandwidth of 50 kHz, for frequencies above 25 MHz, according

to the manufacturer's specifications. Figures 6 to 11 give

our data for the "Wide" bandwidth position and figures 12 to

16 are for "Narrow" bandwidth.

The receiver attenuator was switched to the 40 dB step

in all cases, which is the next-to-most sensitive step. The

signal input level to the receiver was chosen to be 1 mV.

Using the instrumentation of figure 4, with 27 dB L-pad used

alone, the O-to-70 dB step attenuator was adjusted to obtain

a power meter indication of 10 yW. Then the signal was

switched to the input of the 27 dB L-pad and the receiver

gain control adjusted to obtain the correct indication of 1 mV

on the receiver meter.

Figures 6 and 12 show the measured receiver response

vs. signal frequency, in dB above 1 yV. The receiver

selectivity was switched to "Wide" bandwidth for figure 6 and

to "Narrow" bandwidth for figure 12. Two reduced-voltage

bandwidths are indicated on each of these figures, the 3 dB

and 20 dB bandwidths. These represent the frequency width

between points on the curves which have 0.707 or 0.1, respectively, of the maximum voltage response. For frequencies

well removed from the receiver tuned frequency, the "true"

signal level was obtained by the equation:

23
(Signal level, mV) =

r Receiver

li^indicat ion , mV

Receiver noise

^ level, mV

Figures 6 and 12 show two curves, one the actual receiver in

dication and another which has been corrected for receiver

internal noise. A list of the measured values of receiver

internal noise level is given in the following table:

(16)

Rece iver Signal Noise Level of EMC-25 Receiver

Band Frequency

,

Wide Narrow

Number MHz Bandwidth Bandwidth

14 300 0.033 mV 0.018 mV

14 400 0.028 mV 0.015 mV

14 500 0.039 mV 0 .018 mV

15 500 0.0 59 mV (data not taken)

15 600 0.071 mV 0.030 mV

15 700 0.06 7 mV 0.036 mV

15 800 0.062 mV (data lost)

15 900 0.068 mV 0.036 mV

Figure 7 presents experimental data for receiver voltage

response vs. signal frequency, over the range of 299 to 301

MHz. Several types of commonly used bandwidths relating to

field strength and interference meters are indicated on the

curve in this figure. The equivalent -rectangular voltage

passband is shown by the rectangle enclosed in heavy lines.

This bandwidth of 645 kHz was obtained by a numerical integration of the area under the curve of receiver indication

vs. frequency. That is, the area of the rectangle (bandwidth

X maximum indication) has the same value as the area under

the actual plotted bandpass curve. The equation used for

this numerical integration is:

(BW) vol t age

I V(f)

V (f)

m

^

X Af, (17)

where Af is the frequency increment of the s tepped - frequency

signal source. For example, the step size programmed into the

automatic network analyzer was 0.01 MHz in figure 7. The

analyzer was programmed to pause 1 second after phaselock at

each frequency point, before digitizing and recording the

indicated receiver response. Also shown in figure 7 are

several reduced-voltage bandwidths (e.g., 3 dB , 6 dB , etc.)

for a receiver tuned frequency of 300 MHz, as derived from

the volt age -response measurements. These will be referred to

later in this report.

2 4
Figure 8 is derived from the same data given in figure 7,

except that the ordinates are squared values of receiver indication. As mentioned on page 4 , there are two basic types

of broadband signal, one called "impulse interference" and

the other "random noise." The effective random-noise bandwidth, sometimes called "power" bandwidth, is the frequency

interval for which a power gain equal to the gain at mid-band

would transmit the same noise energy as does the actual

power-gain frequency curve. This effective bandwidth can be

measured by dividing the area under the power response curve

by the center frequency value. This is also the random noise

bandwidth, (BW) , obtained from the equivalent rectangular

bandwidth of the voltage-squared passband curve shown in figure 8. The equation used for our numerical integration is:

I [V(f)] 2

[V^(f)]^

It can be seen in figure 8 that the random noise bandwidth is

431 kHz for a receiver frequency of 300 MHz, which is equal

to the 3.33 dB bandwidth of figure 7. The difference between

the 3 dB and random noise bandwidths, in this case, is only

about 5%, having the following ratio:

(BW) ^ (BW) , = 431 kHz v 409 kHz = 1.054. ^ ran ^ 3 dB

Figures 7 and 8 also indicate an "impulse" bandwidth

which is defined later in section 4.5, and has a measured

value [figure 24) of 556 kHz for this receiver frequency. The

difference between the 6 dB bandwidth and impulse bandwidth

was only about 2% in this case. The calculated ratio is

imp ^ CBW)^ = 556 kHz v 556 kHz = 0.98.

Figures 9 to 11 and 13 to 16 are additional graphs of

representative stepped-frequency bandwidth data taken with

the automatic network analyzer. The receiver tuned frequency,

bandwidth setting, band number, etc., are given in the legend

of each figure. Two curves are shown on each figure, using

a solid line for voltage -response data and a dashed line for

squared- volt age - response data. The two equivalent -rectangular

bandwidths are shown by rectangular lines, that is, voltage

bandwidth and random-noise bandwidth. Also indicated on the

figures are the 3 dB , 6 dB and 20 dB bandwidths.

25
299 300 301 302

SIGNAL FREQUENCY, MHz

Fig. 6. Stepped-Frequency Measurement of Receiver Bandwidth

Tuned frequency = 300 MHz (Band 14)

Receiver selectivity = "Wide Bandwidth"

Signal level = 1 mV, Step size = 0.01 MHz.

26
1.1 1

1 1 1 1 1 1 1 1 r

299 .2 .4 .6 .8 300 .2 .4 .6 .8 301

SIGNAL FREQUENCY, MHz

Fig. 7. Stepped-Frequency Measurement of Receiver Bandwidth

Indicated Voltage vs . Frequency

Tuned frequency =30 0 MHz [Band 14)

Receiver selectivity = "Wide Bandwidth"

Signal level = 1 mV, Step size = 0.01 MHz.

27
1 1 1——I

1 1 1 r

SIGNAL FREQUENCY, MHz

Fig. 8. Stepped-Frequency Measurement of Receiver Bandwidth

Squared Voltage vs . Frequency

Tuned frequency = 300 MHz (Band 14)

Receiver selectivity = "Wide Bandwidth"

Signal level = 1 mV, Step size = 0.01 MHz.

28
T

1 1 1 1 1 1 r

SIGNAL FREQUENCY, MHz

Fig. 9. Stepped- Frequency Measurement o£ Receiver Bandwidth

Tuned frequency = 500 MHz (Band 14)

Receiver selectivity = "Wide Bandwidth"

Signal Level = 1 uiV, Step size = 0.01 MHz.

29
SIGNAL FREQUENCY, MHz

10. Stepped- Frequency Measurement of Receiver Bandwidth

Tuned frequency = 700 MHz (Band 15)

Receiver selectivity = "Wide Bandwidth"

Signal level = 1 mV, Step size = 0.015 MHz.

30
o

Q.00

LU

LlJ

1 . 2

1 .1

1 .0

0.9

0.8

0.7

0 . 6

SQUARED

RESPONSE

VOLTAGERESPONSE

VOLTAGE0.3

0.2

[V (f)]

m

6dB BW

RANDOM NOISE ^

BW

25 .40 .70 .85 900 .15 .30

SIGNAL FREQUENCY, MHz

60 900.75

Fig. 11. Stepped- Frequency Measurement of Receiver Bandwidth

Tuned frequency = 900 MHz (Band 15)

Receiver selectivity = "Wide Bandwidth"

Signal level = 1 mV, Step size = 0.015 MHz.

31
.75 299 .25 .50 .75 300 .25 .50 .75 301 .25

SIGNAL FREQUENCY, MHz

Fig. 12. Stepped- Frequency Measurement of Receiver Bandwidth

Tuned frequency = 300 MHz C^and 14)

Receiver selectivity = "Narrow Bandwidth"

Signal level = 1 mV, Step size = 0.0025 MHz.

32
T

1 1 1 1 1 1 1 r

299.75 .80 .85 .90 .95 300 .05 .10 .15 .20 300.25

SIGNAL FREQUENCY, MHz

Fig. 13. Stepped-Frequency Measurement of Receiver Bandwidth

Tuned frequency = 300 MHz (Band 14)

Receiver selectivity = "Narrow Bandwidth"

Signal level = 1 mV, Step size = 0.0025 MHz.

33
I

SIGNAL FREQUENCY, MHz

Fig. 14 Stepped-Frequency Measurement of Receiver Bandwidth

Tuned frequency = 500 MHz (Band 14)

Receiver selectivity = "Narrow Bandwidth"

Signal level = 1 mV , Step size = 0.0025 MHz.

34
1 .

1

o

CO

LU

OC

LU

0.9

0.8

0.7

0.5-

(BW)

ran

699 . 625 .70

[V (f)]

V (f)

m ^ '

3dB BW

RANDOM NOISE BW

VOLTAGE BW

.85 700 .15

SIGNAL FREQUENCY, MHz

.30 700.375

Fig. 15. Stepped-Frequency Measurement of Receiver Bandwidth

Tuned frequency = 700 MHz (Band IS)

Receiver selectivity = "Narrow Bandwidth"

Signal level = 1 mV, Step size = 0.0075 Wlz .

35
Fig. 16. Stepped- Frequency Measurement of Receiver Bandwidth

Tuned frequency = 900 MHz (Band 15)

Receiver selectivity = "Narrow Bandwidth"

Signal level = 1 mV, Step size = 0.0075 MHz.

36
4 . 4 Calibration of receiver impulse bandwidth with a baseband pulse generatoFT

A "standard" impulse generator producing known DC (baseband) pulses was developed at NBS on a previous project [12]

and was available for calibrating impulse bandwidth of the

FIM-2 receiver. This solid-state impulse generator uses a step- recovery diode to generate pulses with a risetime of

about 0.1 ns, amplitude of about 8 volts, and equivalent rectangular time duration of about 0.2 ns . The output spectrum

for this generator is reported as known to ± 1 dB for frequencies up to 1 GHz, and has a nominal amplitude of 2.8

mV/MHz [12]. A plot of the maximum delivered impulse strength

(spectral intensity) vs. frequency is given in figure 17. The

generator is equipped with a built-in 50 -ohm step attenuator

to obtain any desired output below this maximum value, assuming that the load is 50 ohms.

A block diagram of the instrumentation used to calibrate

impulse bandwidth of the receiver is given in figure 18. In

a manner similar to the stepped- frequency measurements of

receiver bandwidth described in section 4.3, the receiver

was initially calibrated for CW voltage at each tuned- frequency

tested. This "voltmeter" calibration was made using a 27 dB

L-pad and power meter, similar to the method shown in figure

4. This initial calibration merely adjusts the receiver gain

control to the proper setting to read correct voltage, as explained in section 4.2. For the data reported here the

signal level used for CW calibration was 1 mV RMS, or

60 dB > 1 yV. The curve of figure 5 was used to set the

power meter to the required value for obtaining a signal of

1 mV.

The principle used to calibrate impulse bandwidth of a receiver is simply to apply a known impulse, expressed in

yV/MHz , and note the receiver indication in yV. The calibrated

bandwidth is then determined from the relation:

Receiver impulse bandwidth, MHz

Receiver indication, yV

Strength of impulse generator, yV/MHz

, or, (19)

RCVR CBW)^^p, MHz = (RCVR Ind, yV)/(I.G. output, yV/MHz)

.

Another way of expressing this same relation, using logarithmic

units, is as follows:

RCVR (BW)j^^p, dB>l MHz=(RCVR Ind, dB>lyV) - (I . G . Output, dB>lyV/MHz)

(20)

37
To illustrate the above equation and to check receiver

linearity, the measured values o£ receiver indication were

plotted vs. applied impulse strength. The data for one case

(receiver tuned to 400 MHz) are tabulated in figure 19 and

plotted as a graph in figure 20. This type of plot is also

used to check for the possibility of receiver saturation

(overloading) during the high-voltage, low-duty-cycle pulses.

The plotted data should be linear, with a slope of unity. That

is, each 5 dB increase in impulse strength should cause a 5 dB

increase in receiver indication. The average value of receiver

impulse bandwidth, at a given frequency, can be obtained from

the best-fit line with 45° slope, as indicated in figure 20.

For example, choosing a signal level of 50 dB above 1 yV/MHz

,

the receiver indication is 43 dB above 1 yV. Therefore the

"average" receiver impulse bandwidth is (43-50) dB or -7 dB

with respect to 1 MHz, which is 447 kHz. The non-linearity

of data is noticeable at both ends of the curves. The departure at the lower end is due to the receiver internal noise,

but the error at the upper end is due to signal compression

in the receiver. The data for the major portions of the curves

indicate that the concept of impulse bandwidth is sound and

usable

.

A summary of all the data for receiver impulse bandwidth,

using the technique of figure 18, is given in figure 21. The

pulse repetition rate of the impulse generator was 300 Hz. As

can be seen, the measured values of impulse bandwidth were in

the range 398 to 544 kHz for the FIM-2 receiver switched to

the wideband pos it ion, with average = 467 kHz and standard

deviation = 53 kHz. This represents a total variation of

2.7 dB for the wideband measurements. The narrowband values

ranged from 68 to 89 kHz, with average = 78 kHz and standard

deviation = 7 kHz. The measured values for specific cases

are compared with other types of bandwidth measurements in

section 6.

4 . 5 Calibration of receiver impulse bandwidth with a pulsed-RF source . A block diagram of another type of instrumentation used

to calibrate impulse bandwidth of the FIM-2 receiver is given

in figure 22. The basis of this standard impulse generator is

the "known" voltage and time duration of the RF burst applied

to the receiver. As shown in the figure, a baseband pulse

generator is used to establish the pulse duration and repetition rate. Each DC pulse overcomes the cutoff bias normally

applied to the solid-state switch by the 10 volt power supply.

In other words, the switch presents a very high impedance to

the RF signal except during the "on" time of the DC pulses.

38
aP *9NI>IMVW IVia 01 NOIiOBMaOO

Oloo id o ld o ldolo

OJi— I— O OO I— 1— CXI CM

I I I I I O

o o o in O Lf> o in O o CO 00 r>» ^

i£) vn

L < ap 'indino xvw 'H19N3illS 3S"indwi

39
SINUSOIDAL

RF SOURCE

POWER = 0 TO 1 W

LOW-PASS

FILTER

25 Ohm —WA

RESISTOR

POWER METER

50 INPUT

(SEE FIG. 4)

48 s^:

2.35

27 dB

L-PAD

BASEBAND BUILT-IN

IMPULSE STEP ATTENUATOR

GENERATOR

50 ^ T-PADS

RECEIVER

NOMINAL Z IN

= 50

I I

I 6" CABLE I h M

Fig. 18. Instrumentation for Calibrating Receiver Impulse

Bandwidth with a Standard Baseband Pulser.

40
RCVR

Bandwidth

Switch

Impul se Generator

Sett in

g

dB>l pV/MHz

Corrected

Imp. Gen.

Output

dB>l mV/MHz

pppp ^ ^^T p-X" dial re ading

,

dB ^1 ijV + 20 dB

Step

+ 40 dB

Step

+ 6 0 dB

Step

+ 80 dB

Step

"Wide" 20 20 . 8 -6.0 - 21 . 0 - 23 . 0 -23.0

25 25 . 8 -0.8 -20 . 0 3 0 30 . 8 3 9 - 14 . 8 u 35 35 . 8 8 . 5 -10 . 4

,, 40 40 8 14 7 - 6 . 5

,, 45 45 . 8 19.2 - 2 . 0 -21. 0 M

5 0 50 8 X 3 . 0 -16 . 1

,,

,, 5 5 55 . 8 X 7 . 9 -10. 6,, 60 60 8 X 13 . 2 - 6 . 8 n 63 63

.

8 X 16 . - 4 . 5 -22.0

u

6 5 65. 8 X 17. 7 -2 . 2 -21.0

67 67 . 8 X 18. 5 -0 . 2 -19.0

M 69 69 8 X 19 . 4 1 . 4 -17.3

"Narrow" 20 20 8 -16.5 -23 . 5 -23. 5 -23. 5 25 25 . 8 -13.9

30 30 8 -11.5 ? 1 35 35 . 8 -7.5

40 40 . 8 -1.8 -20 . 5 45 45 . 8 3.1 -15 . 4 50 50 8 7 . 6 -11 . 8 55 55 . 8 12 .8 -7. 2 60 60 . 8 15 . 9 -2. 5 -21 . 0 63 63. 8 17.1 0 . 2 -17. 8 65 65 . 8 18.1 1

.

7 -15 . 7 67 67 . 8 19 . 2 2 . 5 -14 . 0 69 69 8 20.6 3 . 1 -13. 0

Fig. 19. Check o£ Receiver Linearity for Impulse Interference

Receiver tuned frequency = 400 MHz

Signal source = Baseband impulse generator

Pulse repetition frequency = 300 Hz

41
+80 dB STEP

+60 dB STEP

+40 dB STEP

+60 dB STEP

+40 dB STEP

+Z0 dB STEP

10 20 30 40 50 60 IMPULSE GENERATOR OUTPUT STRENGTH, dB > 1 yV/MHz

Fig. 20 Receiver Linearity for Impulse Interference

(See legend and data in Figure 19)

42
f—

<

+-> (NI OO cn r-~ OO r - -^rHtObOT^-T^OOCn

•H LO cn bO i>- OO OO OO r-- vo OO

is 1 LO ^ to LO

PQ

(I> s in

I—

I

vO ^* to oo c> vO O \0 vO LO \o to o

& A \0 vO LO \0 00 LO Cq to I—1 iH 1—( CNl to I—

1

1—

1

1 1 1 r 1 1 1 (Ni rg (Ni tsi rg (Nj

1 1 1 1 1 1 1

rsi

1 TJ >

u

Pl,

r-i—t 4->

O +-> oo 'i:!- OO O o rsi oo Ti- 00 o •rf o

O rH VO vO «^ LO vo vo ^ LO

LO LO LO LO LO LO LO LO LO LO LO LO LO LO LO LO O A

u

PQ

1—

1

o >

H

•M

oi I—

t

K5 CO o LO «^ VO vO oo oo oo K) iH o

u

Pi H A O CTi O 0~i o •'d- to r:!- -rj- LO ^ LO LO >t:f to to to to to to to to

PI PQ

1—

1

XJ u 5h ,-1 CD o 03 -13 C 03 bO p. bO-H

C PQ

H T3

D rg

O, CO t-H ^

u

PI >

u Id X

CD

5h

PL,

5-1 Cii (D > d XI u 03 e

Cli PQ 3

-H U

> :s ^-J U X! -H

a: c ;2 03 CO

PQ

LOOOOOi—lOOOtO

oa~ioooocTicri

LOOOOOi—lOOOtO

oo^ooooCTicn

oooooooo

tO'^LOLO\ot~^oocr>

oooooooo oooooooo

tOrtLOLOvOt^OOCTl

LO LO

(D Xl

•H 03

o

03 'Z

X3

03

,£3

•H

1=2

CD>

•H

CDU

(D

o

I +-I

l-H 5hCD

<-H CD o u

t/1 CD

(/I M r-t XI 3

•H Ph ^ e

XI ^-l

03 X!

PQ c;03

CD X>

!/) CD iH (/5 ^3 03 Cl-PQ e

I—I nd

X3 03

CD Id p 03 V) +J

03 CO 0

CXI

bO •H

43
Measurements of pulse duration were made with a sampling

oscilloscope. The switch was found to have a 12 ns rise time,

8 ns fall time, and a 90 ns flat -top time, when the baseband

pulser was set to 100 ns . This produces an effective pulse

time, i^g££> of 100 ns or 0.1 ys . Figure 23 shows a tracing

of the RF burst made with a sampl ing -oscilloscope head and

x-y plotter, for the case where the baseband pulser was set

to 70 ns . In this case the measured rise time was 11 ns

(about 5 cycles of a 450 MHz signal), the flat-top time was

52 ns , and the fall time was 7 ns . The calculated effective

time duration for this pulse, as seen in figure 23, is thus

11/2 + 52 + 7/2 = 61 ns . It can be shown that the Fourier

spectrum value, at the pulse carrier frequency, is the same

for the actual pulse and for a rectangular pulse 61 ns long

having zero rise time and zero fall time.

Examination of the oscilloscope trace of the RF pulse

was done both in the coherent mode and in the incoherent

mode, to study the effect of circulators, filters etc. We

were able to produce a "clean" RF pulse with little overshoot,

jitter, or ripple. The term "coherent pulse" is used for one

which is triggered on at a zero crossing of the sine wave and

which contains an integral number of RF cycles. A "circulator"

is a ferromagnetic device which permits energy to flow in one

direction (toward the load) with little loss, but introduces

a large loss to energy flowing in the opposite direction

(toward the source) . Any energy reflected by the switch

during the pulse off-time is dissipated in a 50 resistor.

The oscilloscope was also used to measure a possible difference in amplitude between the steady CW voltage and the RF

voltage during the pulse "on" time. No discernible amplitude

difference was found.

It can be shown mathematically that if an RF pulse contains at least 15 cycles, the calculated worst-case variation

in impulse strength due to non -coherence is approximately 1%.

For an incoherent burst with approximately 10 cycles of RF,

the maximum range is 2%, with an average error of 1%. Thus,

for carrier frequencies above 100 MHz, using an RF pulse of

0.1 ys duration leads to an average amplitude jitter of only

11. For lower carrier frequencies it would be possible to use

longer pulses. Also, it would be advantageous to use coherent

pulses. In this case there would be no uncertainty added to

the calculated value of impulse strength, even for pulses

consisting of only one sine-wave cycle.

44
Similar to the bandwidth measurements described previously, the receiver was initially calibrated for CW voltage with

a 27 dB (or 47 dB) L-pad and power meter. It should be noted

that the measured values of receiver impulse bandwidth, for

the technique shown in figure 22 , are not changed by small

differences in receiver gain, because the receiver is used only

as a reference indicator. The actual measurement of bandwidth

is obtained from the difference of two readings on a step

attenuator, which should therefore be accurately calibrated.

A check was made to determine the effect of pulse repetition frequency (PRF) on the "peak" response readings of the

FIM-2 receiver. The CW signal amplitude was adjusted to

1 mV and the pulse time duration C'^^ff) ^as set to 0.1 ys . The readings obtained on the receiver for a signal frequency

of 400 MHz are given in the following table:

Pulse Repetition Receiver

Frequency Indicat ion

1 kHz 60.0 dB above 1 yV

10 kHz 6 0.2 dB above 1 yV

100 kHz 60.4 dB above 1 yV

1 kHz 60.0 dB above 1 yV

Checks were also made on the effect of pulse repetition

frequency and duty cycle on the measured impulse bandwidth of

the receiver. These tests were done with the receiver tuned

to 300 MHz or 800 MHz. In both cases, changing the PRF from

300 to 1000 Hz decreased the measured impulse bandwidth by

about 10 kHz. Changing the PRF from 300 to 100 Hz increased

the measured impulse bandwidth by about 10 kHz, which is

negligible compared with other system errors.

The experimental procedure for measuring impulse bandwidth of a receiver, using the setup of figure 22, is quite

simple. It consists of noting the receiver indication for a CW signal, turning on the solid-state switch to obtain pulsed

CW, and measuring the ratio between the two values (steady CW

vs. pulsed RF) . The measurement is made by adjusting the 3-

drum step attenuator until the receiver returns to its

original CW indication. The equation used is as follows (see

section 3.9 for definition of terms):

T V (t)

RCVR (BW).^p = \

X (21)

But 1/t = 10 MHz when the applied pulses have ('^)g££ = 0-1 . In other words, as shown in the following sketch, the main lobe

45
of the pulse spectrum has a width o£ 20 MHz (carrier frequency

± 10 MHz) for the chosen RF pulses which have a time duration

of 0.1 ys per pulse. This spectrum is sufficiently broad for

typical receivers having bandwidths less than 0.5 MHz. Therefore the equation used for calibrating the impulse bandwidth

of a receiver, which can then be used for measuring impulse

interference, is as follows:

RCVR (BW).^p, MHz = (10 MHz) '^pk pulsed RF

"

(22)

il

V(f)

380 390 400 410 420

FREQUENCY, MHz

A table of experimental data for measurements of receiver

impulse bandwidth at tuned frequencies between 300 and 900 MHz

is given in figure 24. The CW signal level at the receiver

input terminal (figure 22) was set to 1 mV in all cases. The

FIM-2 receiver was switched to the "Carr" position for CW

signals and to the "Peak" position for pulsed-RF signals. It

can be seen from figure 24 that the measured values of impulse

bandwidth were in the range 537 to 589 kHz for the wideband

measurements, with average = 563 kHz and standard deviation = 16 kHz. This represents a total variation of only 0.8 dB for

all the wideband measurements. The variation when calibrating

the same receiver with a baseband impulse generator (figure 21)

was 2.7 dB . The narrowband measurements (figure 24) range

from 104 to 115 kHz, with average = 112 kHz and standard

deviation = 4 kHz.

An estimate of the maximum possible error in calibrating

the impulse bandwidth of a receiver, using a pulsed-RF generator,

can be obtained by adding the known uncertainties. (a) The RMS

voltage of the CW signal before being pulsed is measured with

a 50 ^2 power meter, which has a calibration error bound of

± II in voltage when used in a 50 ^2 system (e.g., figure 22).

46
BASEBAND PULSE

GENERATOR

T = 0.1. us PRE = 0.1 TO

1 kHz

RF SOURCE.

FREQUENCY = Fo POWER = 0 TO 1 W

CIRCULATOR OR

ISOLATOR,

FOR FREQ Fo

FILTER

40 a 100 n

2.2 uF 1 iiiH

BIAS SUPPLY

+ 10 VOLTS

Y

CONNECT FOR PULSED-RF

DISCONNECT FOR CW

SOL ID-STATE

SWI TCH

H I GH -PASS

& LOW-PASS

FILTERS

STEP ATTENUATOR.

0 TO 70 dB

0.1 dB STEPS

25 Ohm —^WV

RESISTOR

50 Ohm

POWER METER

27 dB OR 47 dB

L-PAD

50 Q INPUT

2.35 Q OUTPUT

RECEIVER

TUNED TO Fo

Fig. 22. Instrumentation for Calibrating Receiver Impulse

Bandwidth with a Pulsed-RF Source.

4 7
FREQUENCY = 450 MHz, PERIOD = 2.22 ns

FALLTIME ^ 3.2 CYCLES ^ RISETIME ^ 5 CYCLES = 11 ns

Fig. 23. Waveform of the Pulsed-RF Burst Measured With a Sampling Oscilloscope.

48
5 M LO rH LO LO o CO rH rH OO

f-> •H r-( r—

1

rH rH rH rH rH rH o rH rH o o

> ;2 rH rH rH rH rH t-H rH rH rH I—

1

rH rH I—

1

H

;2 -

CD (-)

[—( •H O N \o to cn CT) LO CvJ o

:3 > "3 n: LO LO LO LO OO vO l>- OO OO LO H LO LO LO LO LO LO lO LO LO LO LO lO LO

Ba

-

-

+-> >•o o vD 00 rH 00 OO (XI to rH rH to

>-( C r-> '-1 o o U > OO OO OO OO 00 cn cn

+-> •rH u r3 to to to to to to to to to to

03 C 4-J n3 d

o rt OJ - u u to

(Li • 1-1 1—1 •M

+-> o a

1—

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•M

tr, - 0 > •H CD rH rH rH to 1—1 cr, OO o

a W) > PQ M H "Zi LO LO lO LO LO LO LO LO

rv; c rj rsi CM

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<-> 03 I—

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u

H

O

•(->

;-H 03 O rHO

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O

LO

to

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LO

O

LO

LO

to

o o

o o

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to LO

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03 P, +-> I

I r-1

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o c; c; o o

O O O O' o

"=3- 2 2 uo r--

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o c c c o

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03

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r o

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cu +-» > 03 •H i-H U Q

(U

J- ^

1—< CD>

•H Mh -M o u

CD

(/) Mh X

•M W

•H X

;2 +-> C 2

03 CO (L» U

0) !h

(/) :r! rH O

;=! m

!=: KH Cti

1 CO tj cu

(D CU CO

fn (/) ^ rH

(/) a

03 rH

CD S 03 o

(XI

H

49
(b) The effective rectangular pulse duration of the source can

be measured to about ± 31. The error is due mainly to pulse

jitter or non -uniformity in the shape and width of the RF

burst. (c) The spectrum amplitude of the RF pulse decreases

at the edges of the receiver passband, compared with the maximum value Vj^(f) at the CW frequency. This rolloff in amplitude of the spectrum lines, due to the (sin irf x) / (irfT)

envelope, is about 1/2% for a pulse width of 0.1 ys and receiver bandwidth of 1/2 MHz. (d) The error in calibration

of the step attenuator used in figure 22 is about 1%.

(e) Other possible variations in establishing the impulse

strength of the pulsed-CW calibrator (see figure 24) are

estimated to be ± 2%. The overall uncertainty is summarized

in the following table:

(a) Measurement of CW voltage at receiver input terminal = ± 1%

(b) Measurem.ent of pulse duration = ± 3%

(c) Rolloff in amplitude of spectral lines from V (f) = -1/2% ^ ^

(d) Error in calibration of step attenuator = ± 1%

(e) Other uncertainties in impulse strength of the calibrating source = ± 3%.

Maximum total uncertainty = ± 8.51 = ± 0.7 dB.

Root -mean-square uncertainty = ± 21 = ± 0.2 dB.

There is also another error when measuring impulse interference with a calibrated "typical" receiver having a VSWR <_ 1 . 5 , which is connected to a "typical" antenna (or

unknown impulse generator to be calibrated) having a VSWR as high as 2 to 1. This additional possible error due to system

impedance mismatches for the above "typical" case is ± 6 dB

or ± a factor of two for measurement of impulse interference.

It is for this reason that NBS is now developing a new "system"

approach to measuring radiated interference, as mentioned on

page 2 of this report.

5. DISCUSSION OF IMPULSE INTERFERENCE AND

FOURIER TRANSFORMS

There are two possible approaches to devising a standard

of impulse strength (spectral intensity) : (a) design a standard receiver with calculable impulse bandwidth and use it to

calibrate an unknown impulse generator, which in turn is used

to calibrate an unknown receiver, or (b) design a standard

(calculable) impulse generator (source) and use it to calibrate

the impulse bandwidth of an unknown receiver. The first

50
approach requires that the bandpass characteristics of the

standard receiver (narrowband filter) be known mathematically

and that it can be maintained as a stable accurate standard.

The second approach was chosen in our project as being more

direct, fundamental and definable.

Impulse interference, by its nature, has a spectrum

content which covers a wide range of frequencies. It is

assumed that a valid measurand for this type of signal is impulse strength "S", the voltage amplitude per unit of bandwidth.

To simplify measurement of broadband noise, two concepts are

commonly introduced: (a) equivalent rectangular bandwidth of

the receiving system, and (b) impulse strength of the broadband signal. Unfortunately, several definitions of both are

in use. Some of the differences in definition are trivial but

nevertheless inconvenient, for example differing by a factor

of 2 or /2 . Some of the definitions are poorly chosen in that

they are not well grounded in Fourier transform theory. More

seriously, most interference literature de fines impulse bandwidth in terms of the peak value of the receiver response

transient, • Such a definition lacks precision because

it defines the quantity being measured, (BW)

^^^p

, in terms of

the measured response by an instrument (receiver) which does

not have any specified characteristics.

During the past 15 years peak measurements have become

common for measuring impulse interference, particularly by

military agencies and the automobile industry. One reason

for its wide acceptance is the availability of inexpensive

impulse generators which furnish a simple and convenient

procedure for calibrating impulse bandwidth of the measuring

receiver. The impulse generator most commonly found is a baseband (DC) pulser. Mathematically, the "ideal" impulse is

perhaps a delta function, which assumes a DC pulse of infinite

voltage, zero time duration, but known area in volts • seconds

.

The Fourier spectrum of this fictitious pulse has a constant

amplitude and phase for all frequency components from 0 to <». It can be shown mathematically that the impulse strength of

this "instantaneous" impulse in volts per hertz is numerically

equal to double the area of the applied pulse in volt • seconds

.

One good method of describing the response of a receiver

(linear filter network) to a short impulse is in terms of a Fourier transform pair, which then permits description in the

frequency domain. This transform pair is defined mathematically as

51
v(t) = / V(£) e^^""^^ d£ (23)

- oo

00 V(£) = J v(t) e'j^''^^ dt (24)

- oo

where: v(t) = Instantaneous voltage as a function o£ time

V(£) = Transformed voltage, in volt • seconds , as a function of frequency.

In order to deal with quantities which can be handled on

a digital computer it is necessary to use a finite discrete

form of Fourier transform. The coefficients for the complex

exponential form, for uniform time spacing of the voltage

samples, are given by the equations

N-1 j

2ilkn

V (t) = ^ V, (f) e ^ (25) ^ k=0 ^ N-1 -

j

where: v (t) = nth sample of the time wave which consists of

a total of N samples

V^(f) = kth coefficient of the discrete Fourier transform

n=0,l,2,3, N-1

j =

To simplify our mathematical treatment it is instructive

to consider the case of a short rectangular pulse applied to

an "ideal" rectangular filter. This filter (receiver) is

defined as shown in the following sketch, where H(f) is the

receiver ampl itude -vs.- frequency transfer characteristic. That

is, the filter is assumed to have unity gain in the passband

and zero gain elsewhere. Also, it is assumed that the phase

change across the bandwidth is linear, following the relation

({)(f) = (27t) (f^-f) (t^) , where f^ is the center frequency of the

passband and t^ is the transmission time of the filter. In

this case the transfer function is given by H(f) = | H(f) | e^ ^

^"^-^

52
FILTER TRANSMISSION

|H(f )

= - (BW)/?

= fg t (BW)/2

FREQUEUCY, f

PHASE SHIFT FREQUENCY, f

The time -domain response of this ideal filter to rectangu

lar DC pulses can be found by using Fourier methods. First,

by taking the Fourier transform of a rectangular pulse, one

obtains the frequency spectrum as shown in the following

sketch. The mathematical expression is

V(£) = V ^ SiniTfT

(27)

where V = o

T

= Voltage amplitude of DC pulse

Time duration of rectangular pulse, seconds.

53
Multiplying the above expression V(£) for the frequency

spectrum, by the filter frequency response function H(f) shown

previously, one obtains the frequency spectrum of the filter

output. That is, GCf) = V(f) x H(f) . The time -domain response

of the ideal filter output to DC pulses can then be found by

applying the inverse Fourier transform to G(f).

oo vCt) =

J V(f) X |H(f}|eJ'[2^^^ * d£, (28)

Because H(f) is an even function of frequency and (J)Cf) is an

odd function of frequency, this reduces to

00 v(t) = 2 / V(f) X |H(f)| cos [27Tf(t-tQ) * 27rf^t^] df

The limits of the above integral are from f^ to £2 because

|H(f)| is unity between these limits and zero elsewhere.

The above integral can be evaluated approximately if it

is assumed that the value of (sin Trf t) / (irf x) changes so little

in the range f^ to f^ that it may be considered a constant.

This corresponds to the requirement that the filter bandwidth

(BW) is small compared with (1/t) . In other words, for a broadband signal and narrowband receiver, the value of VCf)

is essentially constant over the receiver passband. In this

case the (sin 7TfT)/(TTfT) part of the expression may be moved

outside the integral. Using the trigonometric identity

A - R A +R sin B = 2 sin ) cos (-^) » and the approximation

(f^+f^) - 2f , leads to the final solution:

sin A

vCt) ^ (2V^T)

sin Trf t' 0

r (BW)

irf T

I 0

fsin (7T)(BW)(t-t.))

(7T)(BW)(t-tJ

cos (2TTf^t)

E (2V^T)(BW) (sine Trf^x) [sine (tt) (BW) (t - 1^) ] cos (2Trf^t) .

(30)

Eq. (30) assumes that the receiver gain has been normalized

to unity.

54
FILTER RESPONSE

t 0

v(t)

0

ENVELOPE PROPORTIONAL TO

sinc(TT) (BW) ( t - tp.)

FREQUENCY = f

0

TIME, t

A picture of the expression for v(t) is shown in the sketch

above. As indicated on this sketch, the envelope of the response voltage given by eq. (30) comes from the function

sine (tt) (BW) (t - 1^) . As can be seen from this expression, the

time duration of the receiver response ringing is inversely

proportional to the bandwidth (BW) , because smaller values of

(BW) lead to a slower decay in the envelope -amplitude function.

An oscilloscope trace of the time -domain response given

by eq. (30) can be obtained from the IF output jack of the

usual FIM receiver. This approach to defining impulse bandwidth is described in section 3.2. The maximum value of the

detected envelope of this transient response is the receiver

peak-mode indication. The mathematical value of the peak

reading can be obtained by differentiating the envelope of

v(t) with respect to time, setting this equal to zero, and

then solving the equation for the peak voltage V . This gives

The peak response for this ideal case is seen to be proportional to the strength of the applied impulse (ZV^t) and

to the receiver impulse bandwidth (BW)^^^. If the receiver

is tuned to a frequency f^ which is close to zero, or at

least a frequency which is low compared with the first spectral

zero at f = 1/t, then the function (sine ^^^^1^) is approximately

equal to unity, and

V^(t) = (2V^T)(sinc 7rf^T)(BW).imp * (31)

V^(t) . (2V^t)(BW).imp (32)

55
We were able to obtain greater measurement precision and

repeatibility when using a pulsed-RF generator as a standard

impulse source, compared with a baseband pulser, as mentioned

on page 46, If the impulses applied to a receiver consist

of pulsed RF , the mathematical solution for is very

similar to eq. (32). However, the frequency -domain sketch

for |V(f) I is now symmetrical about the filter tuned frequency

f ^ , rather than being one-sided with maximum amplitude at

zero frequency. This is illustrated by the sketches on

pages 12 and 15. Theory indicates that if the receiver bandwidth is small compared with (1/t), where t is the time duration

of the RF pulse, the peak response of a receiver tuned to f^

is equal to the product of receiver bandwidth and the Fourier

spectrum value at the receiver tuned frequency. That is,

V^(t) = (ViT)(BW).^p. (33)

When a pulse is impressed on a linear filter network,

Fourier methods can be used to analyze the filter response.

For simplicity of analysis it was assumed above (incorrectly)

that the receiver of an interference meter is a linear device.

Also, it should be noted that a single pulse cannot actually

be resolved into a Fourier series. However, the frequency

spectrum of a pulse can be approximated and interpreted by

considering a train of identical repetitive pulses, letting

the period T approach infinity while the time duration per

pulse T remains constant. The amplitudes of the spectral components of the pulses are indicated qualitatively in the following sketches, for progressively smaller values of duty

cycle t/T, where is the voltage of each pulse. It is seen

that for a single rectangular pulse the lines become infinitesimally close together, forming a continuous spectrum

rather than a line spectrum.

56
v{f)|

f, GHz ft

It is interesting to examine the relations between the

"strength" of an impulse, the energy content, and the "pover"

of an impulse source. This can be derived easily for baseband pulses, as follows:

Impulse strength = 2V t, dimensions of vol t • seconds

,

° (34)

2 2 Energy delivered per pulse = = —f-q, joules C35)

Average power delivered = (Energy per pulse)*

(pulses per second)

= Energy/sec

(V^-) (PRF)

50

, watts.

(36)

It is also interesting to

network "stretches" a pulse in

the receiver bandwidth (higher

time required for the receiver

examine how a narrowband filter

the time domain. The narrower

circuit Q) , the longer is the

pulse to decay. That is, the

57
narrower the bandwidth, the longer the receiver remains "inoperative" after each input pulse, due to the so-called IF

ringing. The effect of an interference impulse on a typical

FIM receiver is shown qualitatively in the following sketches

v(t) i

INITIAL IMPULSE —1—

0.2 - us 0.4

IF AMPLIFIER OUTPUT

v(t)

A Ml>l ANTENNA I LIN OUTPUT

v(t)

T <

1 1 ,0.2 0.4

RF AMPLIFIER OUTPUT

v(t)

20 40

ACOUSTICAL OUTPUT

6.

AMP( t)

- ys

0 200

SUMMARY AND CONCLUSIONS

ys

400

The term "bandwidth" is used both for the frequency spectrum of a signal and for the size of "window" in the viewing

receiver. The term "narrowband" or "broadband" interference

is used according to whether the frequency spectrum of the

signal is small or large relative to the bandwidth of the measuring receiver. In order to quantize a broadband signal it is

necessary to use a unit of measurement involving the receiver

bandwidth, such as volts/hertz. Even though two filter circuits may have the same frequency selectivity according to

the customary CW voltage-response -vs . -frequency bandwidth,

they may nevertheless show different responses to the same

broadband interference. It is thus necessary to define socalled effective bandwidths for a receiver.

Broadband interference is generally divided into two

categories, random noise and impulse interference, as discussed

on page 4. Mixtures of the two types can also occur in practice, but we have considered only the ideal cases in this report. Random noise (nonperiodic broadband signal) is best

58
characterized by its power spectral density, that is by the

mean square value of signal passed through the receiver bandwidth. This quantity is defined on page 6. The square root

of random noise is often used as a measurand and is sometimes

labeled as RMS noise level, with the units of volts/Zhert z or

volts per root hertz. Impulse interference (uniform periodic

pulses) is best characterized by its impulse strength, a voltage»time unit ( volt • se conds )

, or by its spectral intensity,

a voltage per unit bandwidth quantity (volts/hertz) . This

quantity may be defined in terms of Fourier transforms as

discussed in section 5.

Several methods for calibrating the so-called impulse

bandwidth of a measuring receiver have been analyzed and checked

at NBS. The approach described in sections 3.9 and 4.5, that

of pulse-modulating a CW carrier, was the best of those we considered. It is theoretically sound, and experimentally it is

simple and direct. The measured data are not as precise as those obtained from a stepped-CW approach, but are more

meaningful for measuring impulse quantities, besides avoiding

the complicated and tedious task of numerical integration.

The type of "known" impulse source chosen in our project

involves measuring the voltage and effective time duration of

a rectangular RF burst, and calculating the impulse strength

from this information. The maximum total uncertainty of this

"standard" source for calibrating receiver impulse bandwidth

was found to be ± 0.7 dB, with a root mean square error of

0.2 dB. We were able to produce a clean RF pulse with 11 ns

risetime, 7 ns falltime, and low ripple. The gating time is

adjustable but we chose an effective pulse length of 0.1 ys

in our experiments, which covered a frequency range of 100 to

1000 MHz.

A table summarizing our measured values of receiver

bandwidth, for various definitions, is given in figure 25.

Most of these data are shown graphically in figure 26. It

can be seen that all the stepped-CW bandwidth curves have a similar shape. That is, if one type of bandwidth is measured

as a function of receiver frequency, for example the 6 dB

bandwidth, then the other bandwidths can be found approximately

by multiplying this value by some factor. The same observation

could be made for the 10 dB and 20 dB bandwidths, which are not

shown in the graph. It can also be seen in figure 26 that the

equivalent -rectangular , vol t age -response bandwidth was always

greater than the 6 dB bandwidth, and that the random-noise

bandwidth was always greater than the 3 dB bandwidth. These

statements hold true for both the wideband and narrowband data.

59
CTS o

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is CQ o o rH rH O O O O o (XI rH O rH rH rH rH rH

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60
-< BAND 14

700 I r

650

600 -

550 -

500 -

450 -

CQ

400 -

350

200

150

100 -

300 400 500 500 600 700 800

RECEIVER TUNED FREQUENCY, MHz

VOLTAGE BW

6 dB BW

PULSED-RF BW

BASEBAND IMP BW

RANDOM BW

3 dB BW

VOLTAGE BW

^PULSED-RF BW

6 dB BW

BASEBAND IMP BW

RANDOM BW

3 dB BW

900

Fig. 26. Measured Bandwidths of the FIM-2 Receiver.

61
The two sets of data for impulse bandwidth, have more

variation than any of the stepped-CW bandwidths. The variability is especially high when using the baseband pulser as a standard impulse calibrator. In fact, the data for the baseband pulser appear to be low and somewhat inconsistent. In

some cases the measured impulse bandwidths, using the baseband

pulser, were less than the random-noise bandwidth. Considering,

therefore, only the data for the pulsed-RF impulse calibrator,

the measured receiver impulse bandwidths were approximately

equal to the 6 dB bandwidth. This is an approximation which

has often been used in practice. For the measurements given

in figures 25 and 26, covering frequencies from 300 to 900

MHz, the average difference between the 6 dB bandwidth and

this impulse bandwidth is approximately 5%. The maximum difference between these two types of bandwidth was 7% for the

wideband measurements (at 900 MHz) and 17% for the narrowband

measurements (at 500 MHz) . . ,

It is instructive to examine the ratios between the

various types of receiver bandwidth measured here, and compare

these with the corresponding values reported elsewhere in the

EMC literature for typical FIM's. The ratios between the

effective bandwidths defined in section 3 depend mainly on the

shape of the receiver selectivity curve. This, in turn, is

determined primarily by the IF amplifier characteristics.

Thus, in general, the ratio (BW) . ^ (BW) is somewhat independent of the receiver tuned frequency. For example, the

calibration charts supplied with our FIM-1 receiver give

graphs for random-noise bandwidth only. A roultiplying factor

of 1.35 is then specified to convert this bandwidth to impulse

bandwidth, for any receiver-tuned frequency of interest. According to the FIM-1 calibration charts, the impulse bandwidth of this receiver can also be obtained by multiplying

the 6 dB bandwidth by 1.0 55. That is,

(BW) . = (BW) X 1.35 = (BW). x 1.055. ^ imp ^ ran ^ ^ 6 dB

According to an example given in "ITEM 1972," page 73 of

reference [8], (BW) = (BW) ^ x 1.07, and

(BW).^p = (BW)^ X 1.093.

The multiplying factors suggested by the HewlettPackard Company for use with their spectrum analyzers [15]

are given as follows

:

62
The peak response to repetitive pulses, according to "HP

Application Note 150-2," has been established and is essentially independent of pulse shape and repetition frequency.

For the synchronously - tuned stages used in their IF amplifier,

the claimed effective impulse bandwidth is 1.5 times the 3 dB

bandwidth. However, several conditions must be satisfied

before this factor can be used, as specified in [15]. They

are: (a) The 3 dB bandwidth must be > (1.7) (PRF), (b) The

product of 3 dB bandwidth and pulse width must be < 0.1,

(c) The peak-pulse amplitude at the mixer input must be less

than that which begins to produce compression, (d) The

receiver must respond to each pulse independently; that is,

the ringing caused by one pulse must decay before the next

pulse occurs, (e) The ratio t/T = duty cycle must be < 0.05,

and (f) The product of PRF and spectrum-analyzer scan time

must be > 10.

A report for the U.S. Navy [2] describes a slightly more

complicated procedure for estimating a receiver's impulse

bandwidth from measurements of voltage -response bandwidth. In

this approach, use is made of a "shape factor" defined as the

ratio of 20 dB bandwidth to 3 dB bandwidth. It is then

necessary to use this "shape factor" to look up a so-called

"correction factor." Finally the receiver impulse bandwidth

and random-noise bandwidth are calculated from the measured

6 dB and 3 dB bandwidths. This approach is demonstrated by

the following example, for the FIM-2 receiver tuned to a frequency of 300 MHz. From the data of figure 25,

MCI, 4^ . „ ^^^^^20 dB 1170 kHz _ ^ "Shape factor" = ^^^^

= 409 kHz "

The "correction factors" for this situation are found from a graph in [2] to be 1.075 for impulse bandwidth and 1.091 for

random-noise bandwidth. Therefore

CB'^^imp = C"fIctor'r)(B"^6 dB ' d'O^SjCSbe kHz) = 608 kHz,

Wran = (TMor' 2" ^ 3 ' (1-091)(409 kHz) = 446 kHz.

This compares with our measured bandwidth values given in

figure 25 as (BW)^^p = 556 kHz (9.4% lower), and

(BW)^^^ = 431 kHz (3.5% lower).

63
Theoretically-derived bandwidth ratios have also been reported in the literature [2] . The conclusions for several

types of filter networks are summarized here as follows:

(a) Rectangular passband, linear phase, (B^)

j^jjjp ' ^^^^ran

(b) Gaussian passband, linear phase, (^W)

^^^p » (/T) (BW)^^

(c) Triangular passband, linear phase, (BW)

^^^^p « (3/2) (BW)^^^

(d) Single resonant tank, actual phase, (BW) * C2)(BW)^^^

(e) Single resonant tank, linear phase, (BW) . » («) (BW)^^

1 mp r an

(f) Cascaded double -tuned IF transformers

No. of (BW)2o ,B (BW)^^^ (BW).^p (BW).^p

^^^g^^ ' ^^^^3 dB ' ^^^)3 dB ' ^^^^ran ' ^^^^6 dB

1' 10 1.5 7 2 1.81

2

~ 4.6 5 1.2 2 1.4 7 1.16

4 3. 38 1.13 1.43 1. 09

• « 2.58 1.07 /I 1.06

From the above table it can be concluded that for most amplifiers used in interference meters, impulse bandwidth is 5 to

10 percent above the 6 dB bandwidth, and random-noise bandwidth

is 10 to 15 percent above the 3 dB bandwidth.

Based on data taken at NBS with a pulsed-RF source, it

is concluded that impulse bandwidth of a receiver is not

related in any simple way to the 6 dB bandwidth or to the

random-noise bandwidth. In all cases, however, the bandwidth

is determined primarily by the IF amplifier characteristics.

The usually-assumed approximation, that of a rectangular

bandpass with linear phase function, is not physically realizable. In fact, this shape of passband is not even desirable

for measurement of impulses. A synchronously-tuned IF amplifier produces a more optimum response. Our experiments showed

that stagger tuning or use of mechanical filters to produce a rectangular passband, will cause severe pulse jitter and

mult iple -response envelopes. A receiver intended for measuring

impulse interference should therefore be designed specifically

for this use. It is not nece ssar ily true that impulse bandwidth is approximately equal to the 6 dB bandwidth.

A summary of several bandwidth ratios measured at NBS

for the FIM-2 receiver, using the data of figure 25, is given

in figure 27. It can be seen from the values in the first

column that the "shape factor" (20 dB bandwidth t 3 dB bandwidth) is much higher for the "narrow" bandwidth (5.67) than

64
it is for the "wide" bandwidth position (2.89). This can

also be observed by comparing the two graphs of figures 6 and

12. The passband "shape factor" and general shape for the

narrowband position do not appear to be a typical or preferred

design for measuring impulses. Considering only the more normal (Gaussian-shaped) curves for the "wide" bandwidth positions, it can be seen that (on the average) the random-noise

bandwidth for this receiver is equal to 1.055 times the 3 dB

bandwidth. The average deviation from the mean value is only

0.51. This average ratio of 1.055 is within a few percent of

the typical design value. The average ratio of (BW)

^^^^ to

(BW)

^ dB ^'^^ wideband data and 1.06 for the

narrowband data. These values are 5 to 10 percent lower than

typically -reported values. It can also be seen in figure 27

that the bandwidth values based on s tepped - frequency CW data

are less variable than those involving measurement of impulse

bandwidth

.

Several possible approaches were given in section 3 for

defining and measuring the various bandwidths of an FIM receiver. If the impulse bandwidth has been calibrated, the

peak value of a receiver response transient can be used to

measure the impulse strength. Impulse bandwidth is a calibrating factor which can be used to convert the receiver indication (yV, peak mode) to the impulse strength (yV/MHz) of

repetitive pulses. It is approximately true that the receiver

indication is proportional to the product of receiver bandwidth and impulse strength. In this case, and for this

reason, it is legitimate and desirable to calibrate the impulse bandwidth of a receiver.

Of the two types of calculable impulse sources evaluated

at NBS (pulsed DC and pulsed RF) , the latter was chosen in our

project, for frequencies between 100 and 1000 MHz. A pulsedRF generator produces a signal which is sufficiently broadband

(much broader than the receiver bandwidth), but has greater

intensity near the rece iver - tuned frequency than a baseband

pulse of the same voltage. This is because the spectral

energy is concentrated at the frequency to which the receiver

is tuned. The maximum value of spectral output for a baseband pulse is at zero frequency (DC) . It is concluded that

the pulsed-RF approach is the most accurate, repeatable and

easy to operate of those we have tested.

65
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66
7. ACKNOWLEDGMENTS

Appreciation is expressed to all those who have contributed time, interest and suggestions for the success of this

project, especially the following members of the tri-service

CCG committee:

J.L. Hayes

S. Feldman

F. Flynn

J.H. McAdams

J. Pyo.

Many of my fellow employees in the Electromagnetics Division of the National Bureau of Standards have encouraged and

helped me in various aspects of this project. Thanks are due

especially to the following persons:

M.G. Arthur -- For technical advice, historical background and being technical reader for this report,

Dr. R.C. Baird -- For his helpful and patient attitude

toward this project.

Dr. P.F. Wacker -- For discussions on Fourier theory and

definitions of terms,

Dr. R.A. Lawton -- For design and testing of the pulsed-RF

impulse standard,

P. A. Simpson -- For measurements and calibrations in the

early stage of this work,

G. R. Reeve -- For useful suggestions and discussions

relating to noise standards,

S.C. Foote and J.R. Becker -- For efficient and accurate

typing and proofreading of this manuscript.

67
8. REFERENCES

[I] D.B. Geselowitz, "Response of ideal radio noise meter to

continuous sine wave, recurrent impulses, and random

noise," IRE Trans. Radio Freq. Interference, Vol. RFI-3,

pp. 2-11, May 1961.

[2] "The radio frequency interference meter," NAVSHIPS

94180, U.S. Navy, 201 pages, July 1962.

[3] F. Haber and R.M. Showers, "Instrumentation for radiointerference measurements," Electronic Industries, pp.

110-118, March 1961.

[4] "An investigation into existing calibration methods of

field intensity meters," RADC -TR- 64 - 52 7 , D.R. Clark,

Project Leader, 228 pages, January 1965.

[5] "Proposed standard for the measurement of impulse strength

and impulse bandwidth," IEEE Subcommittee 27.1, 13 pages.

Draft of April 1971.

[6] M.G. Arthur, "Standards and measurement of impulse

spectral intensity," Unpublished NBS Report, Final Report

on U.S. Army Contract, 56 pages, October 1969.

[7] J.R. Palladino, "A new method for the spectral density

calibration of impulse generators," IEEE Trans., Vol.

EMC-13, pp. 2-7, February 1971.

[8] "Item 1972, a publication of R § B Enterprises," An

interference directory and design guide, 118 pages.

[9] R.B. Andrews, Jr., "An impulse spectral intensity measurement system," IEEE Trans., Vol. IM-15, pp. 299-303,

December 1966.

[10] M.G. Arthur, "Impulse spectral intensity calibration at

the National Bureau of Standards," Proc. 1969 Dept.

Defense Electromag. Compat. Symp . , pp. 133-138.

[II] G.R. Reeve, "Calibration of impulse noise generators,"

Unpublished NBSIR.

[12] A.R. Ondrejka, "Development and evaluation of a solid

state impulse generator," Unpublished NBS Report, Final

Report on Proj . CCG 69-26, 56 pages, December 1970.

[13] "Spectrum Analysis," Hewlett-Packard Application Note 63,

41 pages, August 1968.

68
[14] "Calibration of impulse spectral intensity using pulsed

CW signals," Andy Hish Associates, Tech. Report 0-057010-1,

5 pages, 1971.

[15] "Spectrum analysis ... Pulsed RF," Hewlett-Packard

Application Note 150-2, 33 pages, November 1971.

[16] "EMI measurement procedure," Hewlett-Packard Application

Note 142, 111 pages, July 1972.

69
NBS-1 14A (REV. 7-73)

U.S. DEPT. OF COMM. 1 . PU B L IC ATION OR R K PORT NO. 2. Gov't Accession

BIBLIOGRAPHIC DATA No. SHEET NBSIR 73-335

3. Recipient's Accession No.

4. TITLE AND SUBTITLE

Calibration of Radio Receivers to Measure Broadband

Interference

5. Publication Date September 1973

6. Performing Organization Code

7. AUTHOR(S)

Ezra B, Larsen

8. Performing Organ. Report No.

9. PERFORMING ORGANIZATION NAME AND ADDRESS

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

Boiilder, Colorado 80302

10. Proicct/Task/Work Unit No. 2727281

11. Contract/Grant No.

12. Sponsoring Organization Name and Complete Address (Street, City, State, ZIP)

Calibration Coordination Group

Army/Navy/Air Force

13. Type of Report & Period C ov e red

14. Sponsoring Agency Code

15. SUPPLEMENTARY NOTES

16. ABSTRACT (A 200-word or less factual summary of most si^ificant information. If document includes a significant

bibliography or literature survey, mention it here.) This report covers one phase of a project to improve the accuracy for calibrating fieldstrength meters to measure broadband signals, especially impulse interference. The

amplitude of a narrowband signal is commonly expressed in (J-v, and a broadband signal

is expressed in H-V/MHz bandwidth. A future report (2nd phase) will deal with standards

and techniques for producing a known impulse field to calibrate an antenna-receiver

system in |i.v/(m'MHz), The technique used for initial calibration of a receiver as an

RF voltmeter is new, employing a special L-pad with 48 Q input resistor and 2 Q output resistor across the receiver. The input power across 50 Q is leveled, producing a low -impedance, constant-voltage source at the receiver input terminal.

Broadband interference is generally divided into two types: (a) random noise is best

characterized by its power spectal density, and (b) impulse interference (uniform

periodic pulses) is best quantified by its impulse strength. Several possible approaches

were evalxiated for defining and measuring various bandwidths of a receiver. The re- sponse of a receiver to impulses, as measured with a peak-reading detector, is proportional to the product of receiver impulse bandwidth and signal impulse strength. Of

the two types of impulse sources evaluated here (pulsed DC and pulsed RF), the latter

appears to be the most accurate and repeatable. We were able to produce a clean RF

pulse, choosing an effective duration of 0,1 |jls in our experiments, which covered a frequency range of 100 to 1000 MHz. The worst-case \ancertainty of this standard sourc<

for calibrating receiver impulse bandwidth is i 0.7 dB.

17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons) Broadband interference; field strength meter; impulse

standards; receiver bandwidth calibration; spectral intensity.

18. AVAILABILITY Unlimited

1

' For Official Distribution. Do Not Release to NTIS

1

' Order From Sup. of Doc, U.S. Government Printing Office Washington. D.C. 20402. SD Cat. No. CH

1 1 Order From National Technical Information Service (NTIS)

Springfield, Virginia 22151

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