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NIST Technical Note 1553

NIST Technical Note 1553

An Intra-Volume Ad Hoc Array Concept for

Improved Public-Safety Communications

William F. Young

David W. Matolak

Christopher L. Holloway

Nicolas Bikhazi

Galen Koepke

Helge Fielitz

Electromagnetics Division

National Institute of Standards and Technology

325 Broadway

Boulder, CO 80305

August 2010

""^ATBS O*

'^

U.S. Department of Commerce

Gary Locke, Secretary

National Institute of Standards and Technology

Patrick D. Gallagher Director
Certain commercial entities, equipment, or materials may be

identified in this document in order to describe an experimental

procedure or concept adequately. Such identification is not intended to imply recommendation or endorsement by the

National Institute of Standards and Technology, nor is it intended to imply that the entities, materials, or equipment

are necessarily the best available for the purpose.

National Institute of Standards and Technology Technical Note 1553

Natl. Inst. Stand. Technol. Tech. Note 1553, 60 pages (August 2010)

CODEN: NTNOEF

U.S. Government Printing Office

Washington: 2010

For sale by the Superintendent of Documents, U.S. Government Printing Office

Internet bookstore: gpo.gov Phone: 202-5 1 2- 1 800 Fax: 202-5 1 2-2250

Mail: Stop SSOP, Washington, DC 20402-0001
Contents

Executive Summary iv

1. Introduction 1 2. Signal Optimization and Measurement Methods 2

2.1 Signal Optimization 2

2.2 Spectrum Analyzer and Mechanical Phase Shifters 4

2.2.1 Transmitter and Phase Shifters 4

2.2.2 Receiving Antenna and Spectrum Analyzer 5

2.2.3 Multiple Inputs, Single Output System 5

2.3 Computer with Channel Sampling Cards 7

2.3.1 Single Input, Multiple Output System 7

2.3.2 Transmitter 7

2.3.3 Receiving Antennas and Computer 8

3. Building Structure Descriptions and Experimental Set-up 8

3.1 NIST Building 24, Boulder, Colorado 9

3.2 555 17th Street, Denver, Colorado 9

4. Experimental Results 9

4.1 Spectrum Analyzer Measurement Results 10

4.2 Channel Sampling Card Measurement Results 1

1

4.3 Alternate Representation of Sampling Card Results 13

4.4 Log-distance Path Loss Exponent Estimation by use ofih^Adhoc Array 13

5. Measured Data versus Simulation 14

6. Summary of Results and Conclusion 16

7. References 17

Appendix I: Experiment Setups and Locations 19

Appendix II: Experimental Results 29

111
Executive Summary

This is the seventh in a series of NIST technical notes (TN) on propagation and detection of

radio signals in large building structures (apartment complex, hotel, office buildings, sports

stadium, shopping mall, etc). The first, second, and third NIST Technical Notes (NIST TN 1540,

NIST TN 1541, and NIST TN 1542) in this series described experiments related to radio

propagation in a structure before, during, and after implosion. The next two Technical Notes

(NIST TN 1545 and NIST TN 1546) focused exclusively on RF propagation into large buildings,

with no implosion results. The sixth technical note, TNI 552, focused exclusively on the 750

MHz band due to the availability of spectrum between 764 MHz and 776 MHz for a new publicsafety band. This technical note departs some from the previous six in this series in that here we

present an ad hoc array approach for improving radio-frequency coverage in the public-safety

environment.

The ad hoc array concept was initially introduced in NIST TN 1538 in the form of a boundary

condition and optimization analysis problem. The term ad hoc array refers to the use of several

small communication devices that are randomly placed in an area and hence create a real-time

communication network. In large, complex building structures, the signal-to-noise ratio often

drops below a sufficient level for establishing a communication link between the transmitter and

receiver, thus leaving personnel without radio coverage in sections of the building. This ad hoc

array concept makes use of the wireless ad hoc nodes (a node refers to one communication

device) as elements of an array to coherently phase the signals at the receive location (or

coherently phase the multiple received signals from a transmitter in the reciprocal process), to increase the signal-to-noise ratio. The single transceiver is typically located in the far-field

radiation pattern of the individual array nodes, but within the volume that contains the complete

set of ad hoc array nodes. Hence, we describe the system as an intra-volume ad hoc array.

Simulation results from the initial work and related publications demonstrate the potential

benefits of this ad hoc array concept. Results provided in this report cover representative realworld measurements of the ad hoc array by use of two different measurement techniques. These

measurements focus on the frequencies of 750 MHz and 2.4 GHz, and are based on using the

array in a receiving and transmitting mode, respectively. The two representative environments were a large basement room with a connected hallway and stairwells, and an atrium between two

large office buildings in an urban setting. Key findings are that the ad hoc array concept

provided gain performance as predicted from previous analysis, simulations, and laboratory

experiments. The amount of measured gain ranged from 2 dB to 6 dB for an array of two

elements, and up to 10 dB for a four element array. The two element results suggest that using even only two wireless nodes in a coordinated manner can provide significant improvement for

public-safety wireless communication systems.

While the primary focus of this effort was to measure the real-world performance of an ad hoc

array concept, we also provide a different mathematical framework than the original

electromagnetic optimization analysis to assist communication engineers in applying the

proposed approach to wireless systems in a real-world setting. This combination of measurement

results and intuitive mathematical framework serves as the basis from which to design a wireless

ad hoc array system. The natural next step in the process is to develop wireless nodes and

protocols that can implement the ad hoc array technique.

IV
An Intra-Volume Ad Hoc Array Concept for Improved Public-Safety

Communications

William F. Young, David W. Matolak, Christopher L. Holloway, Nicolas Bikhazi,

Galen Koepke, Helge Fiehtz

Electromagnetics Division

National Institute of Standards and Technology

325 Broadway, Boulder, CO 80305

Here we report on the testing and measurements of an intra-volume ad hoc array concept

suitablefor public-safety communications in buildings. The term ad hoc array refers to the use of

several small communication devices that are randomly placed in an area and hence create a

real-time communication network. The overall concept is to use randomly located or placed

wireless devices in a coordinated manner in order to increase the radio-frequency signal at

otherwise unreachable locations. A typical ad hoc network is limited to a coverage area

achieved by the useable coverage of single nodes. We seek to extend the radio-frequency

coverage by using two or more nodes (a node refers to one communication device) as elements

of an ad hoc phased array. Previous simulations and theoretical analyses have suggested the

benefits of the array concept. The measurements presented here, collected in real-world

environments, demonstrate that the ad hoc array technique can provide useful gain, up to 10 dB

with onlyfour elements. The measurements also indicate a typical gain of2 dB to 6 dB by use of

only two elements.

Key words: ad hoc phased array; emergency responders; multipath; excess path loss; publicsafety communications; radio communications; radio propagation experiments; weak-signal

detection; wireless array

.

1. Introduction

When emergency responders enter large structures (apartment and office buildings, sports

stadiums, stores, malls, hotels, convention centers, warehouses, etc.), radio communication to individuals on the outside is often impaired. Mobile-radio and cell-phone signal strength is reduced due to attenuation caused by propagation through the building materials and scattering

by the building structural members [1-8]. In addition, a large amount of signal variability may

be encountered due to multipath reflections throughout the structures, which can cause severe

signal degradation in communication systems.

Here, we report on a project conducted by the National Institute of Standards and Technology

(NIST) to investigate the concept of a wireless ad hoc array to help mitigate the communications

problems faced by public-safety personnel (firefighters, police and emergency medical

personnel) in disaster situations involving large building structures. The project was sponsored

by the Department of Justice Community Oriented Policing Services (COPS) program through
the NIST Public Safety Communications Research Lab (PSCL). This project focuses on two

frequencies, 750 MHz and 2.4 GHz. The 750 MHz band is currently of interest because the FCC

is in the process of allocating spectrum between 764 MHz and 776 MHz for a nationwide, nextgeneration interoperable broadband network for use by the public safety community. The

2.4 GHz band is in a widely used Industrial, Scientific, and Medical (ISM) band.

A possible option to improve radio-frequency coverage inside large buildings is to use ad hoc

netv\'orks that provide node-by-node connectivity (i.e., communication to each node) between the

emergency responders and on-site communication facilities. However, these traditional ad hoc

networks can communicate only within the range of a single node, which means that failure of a

single node or insufficient node transmit power, results in a failure of the communication link.

This ad hoc array actually utilizes several coordinated nodes, thus removing or at least

significantly reducing the dependency on any single node. The ad hoc array concept could

actually be applied to maintain connectivity in a traditional ad hoc network if the network

experiences segmentation.

An important point about the wireless ad hoc array concept tested here is that it is neither a

traditional ad hoc wireless network nor a traditional phased antenna array. In a traditional

wireless ad hoc network, the process is to establish node-by-node connectivity. In that case, the

transmission range is limited to the range of a single node. A traditional phased-array (when

operating in transmit mode) assumes the receiving location is in the far-field of the complete

array volume. In our wireless ad hoc array, the receiving node is within the volume of the array. Our approach represents a merging of the wireless ad hoc network and phased array concepts.

The experiments reported on here took place in two settings, the basement of a two story

laboratory/office building and the atrium between two large buildings. Two different systems were used to test the ad hoc array concept in receive and transmit modes. A basic description of

the wireless ad hoc array concept, and a description of the two systems used to test the concept

are discussed below.

2. Signal Optimization and Measurement Methods

2.1. Signal Optimization

Use of randomly located wireless devices in an intelligent and coordinated fashion for

emergency responders was initially investigated in [9-12]. These initial efforts started with a representation of the optimization problem as infinitesimal dipoles in the presence of several

boundary configurations, and used simulation studies to investigate the potential gain available.

Optimization means the selection of complex weightings, i.e., phase and amplitude, so that the

transmitted signals from the array nodes arrive co-phased at the receive node, with amplitude

levels that optimize the aggregate power of the transmitting nodes. In other words, each array

node is multiplied by a complex weighting so that the relative amplitudes amongst the nodes are chosen such that those nodes with more power available with respect to the receive location

provide the greater contribution. If the environment is free space, then the array nodes closest in

Euclidean distance (i.e., direct path) to the receive node provide the most power. However, in a building environment, the closest nodes may not provide the strongest contribution, due to reflections and scattering off furniture, walls, doors, windows, etc. Thus, the complex weights
must account for the effects of the physical environment. References [11], [12] include results

from the supporting laboratory experiments that validated the simulation findings. This report covers the next step in the proof-of-concept process, where the testing is extended to real-world

settings.

In the optimization, both the power level and the relative phases between the wireless elements

are controlled. However, as both the simulation and previous laboratory experiments

demonstrate, adjusting of the phase alone accounts for much of the benefit. This simplifies the

procedure that devices would need to implement. Thus, here we will use a co-phasing approach,

where the individual phases of the ad hoc array elements are adjusted so as to add in-phase at the

receive location when the array elements are transmitters. (Co-phasing means adjusting the

transmitted signal phases of multiple transmit elements such that all the received signals add

coherently—see [13], [14].)

The array is either operated in receive or transmit mode. In the receive mode, there is a single

transmitter, and the received signals from individual array nodes are co-phased before addition,

creating a single received signal. In the transmit mode, each array node is transmitting a signal

with additional phase adjustment that ensures co-phasing of all the array elements at a single

receive location. In the context of current communication system terminology, the case of a transmit array sending to a single receiver is a multiple-input/single-output (MISO) system,

whereas the case of a single transmitter sending to multiple receivers is a single-input/multipleoutput (SIMO) system.

The simuladon results in our earlier work [9-12] point out some key considerations in

construcdng a field-deployable implementation. First, diminishing returns in total received

power occur once the array contains a small number of transmitters, approximately four

transmitters for the opdmized case (i.e., both amplitude and phase weighting) and six to eight

transmitters for the co-phased case (i.e., only phase weighting). Second, for this same small

number of transmitters, the modified directivity results are not significantly different for the

optimized and co-phased results [9], [10], [12]. Thus, we propose the following algorithm to obtain the co-phased results, with the recognidon that the algorithm also provides a reasonable

approximadon of the opdmized modified directivity. Note that this algorithm pertains to the

receive mode operation, and a reciprocal process applies to the transmit mode.

1) Turn on each transmitter individually, and rank the received power of each transmitter.

2) Select the strongest contributing transmitter, and turn it on with all the others off. 3) Select the next strongest transmitter determined in Step 1, and turn it on.

a. Adjust the phase of this transmitter through 360°, and determine the phase when

the peak power occurs at the receiver. (Note that in these experiments, we are using a mechanical hand-tuned phase shifter. However, previous simulations

indicate that stepping at 22.5° will allow the co-phasing process to achieve very near the maximum. We demonstrate this in the receive mode system discussed

below by applying a four-bit digital phase shifter to the signals.)

b. Set the phase of this transmitter to that corresponding to the maximum pow cr at the receiver.
4) Repeat Step 3 for the remaining transmitters (with all previous transmitters on and

adjusted), until the desired number of contributing transmitters is reached.

The measurement results here are obtained from the co-phasing approach. An important

observation made in [11], [12] is that the co-phasing process is not highly sensitive to the phase

deviation. In other words, near maximum gain results are achieved by co-phasing signals within

± 22.5° of each other or a 45° spread.

This process will ensure co-phased transmitted signals at the receiver (for receive mode

operations) for however many transmitters are used in the array. If total system or radiated power

is not a limitation, all the co-phased transmitters could be included in the process. However, if system power efficiency is important, say for example in a low-power sensor network, then

transmit power levels of nodes are weighted such that nodes able to provide more power to the

receive location contribute more power, while weaker node contributions are reduced. In free

space with omnidirectional radiation, this simply means the power contribution is inversely

related to the geometric distance between the array node and the receive location. This

combination of selecting power levels and phases is called the complete optimization or the

optimized approach, since it optimizes the overall system power; i.e., the aggregate of the array

node powers, while ensuring co-phased signals from the array nodes at the receive locations. In an environment such as a building, the optimization process still applies, but the optimal

individual power settings are not necessarily inversely related to separation distances.

2.2. Spectrum Analyzer and Mechanical Phase Shifters

2.2.1 Transmitter and Phase Shifters.

The ad hoc array was composed of a signal generator that fed radiating antenna elements

composing the ad hoc array. A 1:8 power divider split the power evenly across the array

elements, and mechanical phase shifters allowed adjusting of the phase. The mechanical phase

shifters cover 360° at 2.0 GHz, and greater than 360° of phase shift at 2.4 GHz. However, at 750 MHz, the amount of phase shift is less than 150° so this measurement setup was used only

for the 2.4 GHz case. Figure 1 shows a diagram of this setup, and Figure 2 displays the various

components of the measurement system.

A method of selecting array antenna elements was provided by electro-mechanical relays and

selector switches depicted in Figure 2. The output from each phase shifter was either fed into an

antenna element via a 36.6 m RF cable or terminated in a 50 O load. (The cable loss was

approximately 20 dB.) One of the eight output ports from the power divider was terminated in a 50 Q load connected to a phase shifter for all measurements, because only seven relays were

available. Thus, only seven array elements were used in these experiments. Monopoles mounted

on aluminum octagon ground planes with an approximate center-to-vertex measurement of

0.24 m were used for the 2.4 GHz antennas. The ground planes were supported by 1.9 cm

{V^ inch) diameter polyvinyl chloride (PVC) stands at a height of approximately 1.3 m. Figures 3

and 4 show several of the ad hoc array elements at locations in the Building 24 basement and the

building in downtown Denver, respectively
2.2.2 Receiving Antenna and Spectrum Analyzer

The performance of the ad hoc array was measured by placing a receive antenna either within the

volume of the array or in the vicinity of the array boundary. The receiving antenna, identical to the transmitting array element antennas, was connected to a spectrum analyzer via an 18.3 m RF

cable, and the measured value was obtained by reading the value displayed on the spectrum

analyzer. Depending on how rapidly the channel changed and on the level of the received signal

above the noise floor, the waveform displayed on the analyzer exhibited variable rates of

fluctuation. However, because the intent of these experiments is not to obtain a precise

measurement of the absolute received signal power, but rather to quantify the amount of gain

possible by applying the ad hoc concept, the precision of visual observation and recording of the

power spectrum was deemed adequate. Consistency of results from this manual approach with

the automated data capturing system described below demonstrates that the manual measurement

approach provides sufficient accuracy.

2.2.3 Multiple Input, Single Output System

The spectrum analyzer and mechanical phase shifters constitute a multiple-input, single-output

(MISO) system. We provide a mathematical description of this MISO system, followed by some

implementation particulars of the transmitting and receiving components of the system. The

mathematical description here differs from the optimization analysis based on approximate

boundary conditions used in the earlier works [9-12], and is intended to provide a more intuitive

description for communication system engineers. The development here follows from the

information theory perspective. A good discussion on MISO, and single-input, multiple-output

(SIMO) systems in wireless channels is found in [15].

In the MISO field tests, N transmit antennas were used to send a signal to one receive antenna,

{N=l for the MISO system tests). The source transmit signal was

s{t) = Acos{coJ), (1)

where, without loss of generality, we let the phase of the sinusoid be zero. The fi"equency used

wasyc=^c/27i=2.4 GHz. The input signal has average power Ps=yl^/(2Z) (in watts '\f A is in volts

and the impedance is Z in ohms); as is conventional, for simplicity of signal analysis we let Z=l Q. This signal was split approximately equally in power and connected to the N transmit

antennas. Connection was via a power splitter followed by an adjustable phase shifter on each of

the A'^ branches. Thus each transmit antenna had an input power of P'Jn -{a\ /(2Z) where the

primed values of P and/i include all losses associated with the power splitter, phase shifters, and

the connecting cables. Each transmit antenna experiences nearly equal losses due to the power

splitter, phase shifter, and connecting cable. The signal input to the /"^ antenna can be expressed

as
where A. = A / yJlN = constant, and the phases {^^,}^i are due to spUtter, cable, and connector

delays, and are nearly identical for all branches. Note that for these tests, the key conditions are

that y/. is constant and that s. (amplitude and phase) is approximately equal for all the transmit

antennas. The phases {^,}^i are the adjustable phases set by the phase shifters, and are used to remove the phase differences due to the propagation paths between transmit and receive antennas

in order to achieve the maximum received signal at the receiver (This is evident after convolving

with the channel impulse response, as discussed below). Each signal of the form in (2), when

output from the antenna, incurs a gain scaling and a phase shift as a function of elevation and

azimuth angles. For any given elevation and azimuth angle, we can incorporate the phase shift

into the constant phase ^, and can similarly incorporate the gain into A.

.

Measurements were taken at two different sites, one on the NIST campus in Boulder, CO and the

other in downtown Denver, CO. The measurements at the NIST site in Boulder, CO contained a very limited amount of object movement within the array area, i.e., people, and all the receive

antenna test locations were indoors. At the Denver site, there was sporadic pedestrian motion in

the environment, and vehicular motion outside on the streets. The receive antenna was indoors

for all test cases except the first location. Thus, for simplicity we assume the channel is time

invariant. For the sinusoidal signals of (2), we can employ the bandpass channel impulse

response (CIR), which is given by [16],

/2,.(r) = X«,,^(r-rJ, (3)

A-=l

where / indexes the channel from the f'^ transmit antenna to the receiver, Li is the number of

multipath components in the /^ CIR, and the amplitude of the k^'^ multipath component (MPC) in

the i'^ CIR is aki. The J is a Dirac delta function, and Tki represents the delay of the k' MPC of

the i''' CIR. We can employ values for the MPC amplitudes from measurements, but these will be

only approximate. Generally, models consider the a's to be drawn from a random sample with

statistics specified by a large number of measurements.

Because the channel is linear, the output of the receive antenna due to the i''' transmit signal is given by the convolution of (2) and (3); i.e.

r, (/) = s, (t) * h, (r) = V^^Z^^' ^os[6>, (t - r,, ) + ^, + ^. ]

(4)

k=l

where 6ki=\i/i - cOcZk can be considered the aggregate channel phase term, which can also

incorporate the receive antenna phase. The total signal at the receiver is the sum of the A^

transmitted signals:

*
KO = Zr, (0 - ^, ZZ^<^ ^os(^,/ + ^,, + ^, ). (5)

/=1 /=1 A=l

The link distances varied from approximately 3 m to 50 m or more. The channel phase terms are ^^-27ix2.4xl0^%for the frequency used for spectrum analyzer system. For these link distances,

line-of-sight values of tki range from approximately 10 ns to 150 ns, and are even larger for

reflected signals. Using this range of values of delay, the channel phase terms span the range of

many multiples of 2;7-. Thus representing these phases {Oki) as random and uniformly distributed on the range [0, In), or Q<6ki<2n, is a very good model. The phase shifter phases {^,}f=i were

adjusted to maximize \r(t)\.

While this report focuses on the measured performance of the ad hoc array, several simulation

results are included to demonstrate the usefulness of analyzing the ad hoc array performance

with the mathematical framework presented above. The simulation discussion follows the

experimental results in Section 4.

2.3. Computer with Channel Sampling Cards

2.3.1 Single Input, Multiple Output System

The first method of data collection constituted a MISO system. However, the reciprocal

approach, namely using one transmitter and multiple receiving nodes in an ad hoc array manner,

i.e., a SIMO system, can also provide similar signal gains. We used a SIMO system discussed

below to demonstrate the bi-directional benefit of an ad hoc array communication system.

The SIMO system used in the field tests supported automated data capturing with the ad hoc

array operating in receive mode. In this set of tests, a single transmit antenna was used to send a

signal to A'^ receive antennas, where N= 1, 2... 6. The transmitted signal is in the same form as

(1), but in this case the frequency wasy^.=750 MHz. Here we have A^ individual received signals,

each of the form of (4). Each received signal was down-converted and sampled in a separate

hardware channel. The individual received signals were then combined at baseband to maximize

the amplitude \r(t)\ via adjustment of the phases, analogous to the procedure for the MISO tests.

The complex baseband equivalent of (5) is m= 4 ZZ^-A (0 exp[7(^„ (0 + % )] • (6)

/=1 k=\

The amplitudes and channel phases in (6) are actually slowly-varying functions of time due to the (slowly) time-varying behavior of the channel. Thus, the time dependency is now explicitK

shown in the aik(t) and Oki(t) variables.

2.3.2 Transmitter

The fixed transmitter setup included a signal generator outputting a single fi^equency at 750 MHz, an 18.5 m cable, and a loaded dipole antenna mounted on an akiminum octagon
ground plane with an approximate center-to-vertex measurement of 0.24 m. This same ground

plane is used for the 2.4 GHz antennas. The output power on the signal generator was set to 20 dBm, the cable loss was approximately 6 dB, and the gain of the anterma approximately

3 dBi. The antenna and the ground plane were supported by a PVC stand at a height of

approximately 1.3 m. Figures 5 and 6 show the 750 MHz setup and data collection system and

Figures 7 and 8 show some of the 750 MHz antennas at the Building 24 and Denver experiment

sites, respectively.

For one experiment in the Denver building, a handheld 750 MHz transmitter was carried

throughout the array volume and perimeter. The transmitter power was set for an output of

30 dBm (instead of the 20 dBm as above), and the "rubber duck" anterma had a gain of

approximately dBi (note that this is a different anterma and transmitter from that used for all the other experiments.).

2.3.3 Receiving Antennas and Computer

This automated data capture setup utilized computer cards that sampled six channels directly at 750 MHz, converted the sampled channels to baseband, and then performed the co-phasing

algorithm on the six down-converted charmels. A raw sample rate of 1 ns was used to capture

8192 instances of the channel samples, for a total capture period of 8.192 )is. These 8192

samples were converted to baseband, and averaged to obtain a single measurement of the

received signal, which was then used in the co-phasing algorithm. The overall sampling rate for

the processed data was approximately 1.7 samples per second. In this case, a single transmitter

provided the signal sampled by the six channels. Each of the six channels was cormected via a 36.7 m RF cable to one element of the ad hoc array. Each element consisted of an anterma,

ground plane, and PVC stand identical to the fixed transmitter setup. An important feature of this

automated system is that the channels are sampled simultaneously. Thus, the physical

environment is the same for each of six array elements when a particular sample is collected.

This process represents the reciprocal (or "dual") process to use of multiple transmitting

elements and a single receive element. In this approach, there are six receive elements and a

single transmitter. Changes over time in the physical environment, e.g., pedestrians walking by,

and the difference in frequencies i.e., 750 MHz versus 2.4 GHz, mean that the channels may be

different. However, the physical locations of the array elements are the same in both cases, and

the same co-phasing algorithm is applied to the received signals. Due to frequency limitations of

the cards, this process was performed only at 750 MHz. The specific cards used in the

experiments are the CompuScope 21G8, and the data sheets are available at [17]. The computer

was a Dell Precision 380 running Windows XP, with a Pentium D 3.0 GHz processor and 2 GB

of memory. Mention of any company names serves only for identification, and neither

constitutes nor implies endorsement of such a company or of its product by NIST.

3. Building Structure Descriptions and Experimental Set-up

This section briefly describes the two building structures used in these experiments, and the

experimental setup. While the locations of the ad hoc array were essentially random, the general

locations of elements were chosen to test possible field-deployment configurations. For example,

two of the array elements were placed in a hallway off the main room in the NIST Building 24
experiments, while three array elements were placed outside of the atrium at the Denver

experiment. Both cases represent possible deployment configurations of an ad hoc array system

by emergency responders. More specifics on the two experimental sites follow below.

3.1 NIST Building 24, Boulder, Colorado

Building 24 on the NIST, Boulder campus is a combination laboratory and office structure. The

structure is composed of typical building materials, including concrete, steel, drywall, and

acoustic ceiling tiles. The section used for the array tests consisted of a large, open room in the

basement, a connected hallway on the same level, landings in two stairwells that connected the

basement to the upper floors, and a small bathroom at the basement level. Figure 9 shows the

layout of the ad hoc array elements and the test locations. Test locations represent either a receive antenna connected to the spectrum analyzer for the 2.4 GHz measurements or a transmit

antenna connected to the signal generator for the 750 MHz measurements. Note that array

element E is not used in the sampling cards and computer data collection, as the system has only

six available channels.

Three important features of the array layout and test locations are: (1) the placement of two array

elements in the hallway, which represents the case when array nodes are located in the hallway

of a building, (2) the inclusion of a test location within a bathroom, which represents the case when an array node is located in a separate room, and (3) the location of an array element and

test location on stairwell landings approximately 1.3 m above the basement floor level, which

represents the case when the array and single receiving/transmitting node are located at different

elevations in the building. Figure 3 and Figure 7 show some of the basement features, including

the concrete block walls and the unfinished ceiling.

3.2 555 17th Street, Denver, Colorado

The 555 17' Street location is an atrium or enclosed lobby area between two buildings in

downtown Denver. (We refer to the location as the Denver site.) The structural construction

materials are a typical high-rise combination of concrete, glass, and steel, and the lobby area

consist primarily of stone, glass, and metal frames. The layout of the ad hoc array elements and

the test locations both within the atrium and on the sidewalk adjacent to the building are shown

in Figure 10.

Two key features of this layout are: (1) the placement of the three array elements. A, B, and C, on the sidewalk outside of the atrium area, and (2) the location of test point 1 outside the main

entrance of the atrium. This is to emulate the use of the array concept to help with signal

penetration into the building. The experiments in Denver also include a scenario in the 750 MHz

test where a handheld transmitter is carried on an arbitrary path along the sidewalk where A. B.

and C are located, and throughout the atrium.

4. Experimental Results

The experimental results for Building 24 and the Denver site are provided for both measurement

systems in the layouts and test locations described previously. As is evident in the data plots, the
computer system with the network cards allows the collection of a much greater amount of data over a given period. However, overall results show consistent behavior between the two

approaches, and both validate the ad hoc array concept.

The expected combined systematic and random errors introduced by our measurement

equipment are expected to be on the order of tenths of decibels. Two different data collection

methods were used for the two different frequencies. For the 750 MHz test, an automated

system was used, and either 50 or 100 samples were collected at each test position. The

repeatability for these 50 to 100 samples were within 0.5 dB. This variability is due mainly to the

propagation channel changing during the experiments caused by objects moving during the

experiments. While no repeatability data were obtained for the 2.4 GHz measurement, the

uncertainties in both this set of experiment are believed to be less than IdB.

4.1 Spectrum Analyzer Measurement Results

Figures 11 to 16 show the application of the co-phasing algorithm to the 2.4 GHz case, or the

transmit mode. The first set of figures. Figures 1 1 and 12, show the raw results of the co-phasing

process. In the Building 24 case, from one to seven transmitting elements were used for all eight

of the receiving or test locations. At the Denver site, from one to seven transmitting elements were used for five of the test locations, and from one to six transmitting elements were used for

the remaining two test locations (test locations 3 and 7). In the case of test location 3, array

element G was not included, due to the close proximity to avoid dominating overall performance

by a single element (G). At test location 7, array element D was not included for the same

reason. Both of these cases represent conditions where the array concept would be unnecessary

since there is a very strong signal emanating from the nearby node.

While the increase in recorded power provided by adding co-phased elements is evident in the

raw data, due to the difference in distances from array elements at various test locations, the

relative increase is not obvious. Figures 13 and 14 display the relative increase by normalizing

the data with respect to the strongest contributing element. Thus, the initial single element starts at dB for all test positions. The results for Building 24 in Figure 13 indicate an increase of 1 dB

to 6 dB when the second array element is included. The 6 dB increase is at the theoretical upper

bound, assuming identical gain patterns from the antennas at the test location. In other words,

assuming that each array element is capable of establishing the same electric field level at the test location, with two elements, at best, the electric field magnitude can be doubled. This

corresponds to a 6 dB increase in power.

For the Denver site results shown in Figure 14 for test locations 4 and 6, the gain from two

elements is 6.3 dB and 6.1 dB, respectively. This would appear to be a violation of the

theoretical maximum; however, that maximum does not take the environment into account. In the case of test location 4, there is a column near the receiving antenna, which affects the antenna

gain pattern. Similarly, many of the array elements are near objects that affect the antenna gain

patterns. Thus, we do not expect identical performance from each array element. In addition, if the test location resides in a deep null, then any slight change in the physical environment (i.e., pedestrian movement, door opening, etc.), can significantly change the measured power.

10
One of the important factors for a practical system is determining the number of array elements

to utilize. Figure 13 indicates that the gain increase from one to four elements ranges between

2 dB to 10 dB. However, the increase from four to seven elements only provides an increase of

approximately 1 dB (which is approaching the measurement uncertainty). In Figure 14, the gain

increase from one to four elements ranges between 5.5 and 10 dB. Similar to the results in Figure

13, the increase of four to six or seven elements provides only about a 1 dB increase. Thus, while

the Building 24 case shows a wider range of gain at four array elements, both the Building 24

and the Denver site results show very similar behaviors on the upper limit of four elements and

in the trend from four to seven elements (Note that the theoretical upper limit for normalized

power in free space is an N^ curve. Thus, for four elements the maximum theoretical power gain

would be 12 dB, and 16.9 dB for seven elements, assuming a free-space environment and equal

contribution from the elements at the test location).

A key aspect of a practical system is the ability to increase a weak signal to a level that supports

communication. Clearly, if the signal is sufficiently strong already there is minimal value to the

ad hoc concept. However, the array concept can help raise a weak signal to a useable level. Another important aspect of the array concept is the ability to extend the perimeter or boundary

of communication connectivity. Figure 1 5 and Figure 1 6 demonstrate that two elements can be

used to (1) raise the signal to a useable level, and (2) extend the communications boundary.

In Figure 15, the two elements located in the hallway off the main room in the basement are co- phased to extend the communication boundary. If one of the signals is well above the noise

floor, the addition of another weak signal provides little benefit; e.g., see receive test position 3

in Figure 15. However, receive test positions 7 and 8 both demonstrate how using two elements can provide substantial gain in the signal. As discussed above, there initially appears to be a violation of the 6 dB maximum possible at receive position 8. This is likely due to not starting

from the absolute maximum for each individual element; i.e., not adjusting the phase to account

for the multipath created by a single element. It is interesting to note that, other than receive test locations 3 and 6, all receive test locations indicate a gain of at least 2 dB by use of these two

elements in the hallway.

Figure 16 shows the results for two array elements at the Denver site. In this case, receive test position 4 provides a gain of almost 15 dB. This better than expected maximum behavior is likely due to not establishing the individual maximum for each of the two elements before

optimizing the combination of the two. In other words, the two elements may each have been

creating a deep self-null due to several multipath contributions, while the co-phasing process in the phase adjustment removes the self-null and maximizes the contribution of the second

element. However, more importantly, across all receive positions, at least 1 dB of gain was

realized, and in most cases, more than 2 dB was possible when two relatively weak signals were

co-phased.

4.2 Channel Sampling Card Measurement Results

The data collection and subsequent application of the co-phasing algorithm to the 750 MHz

signals provides results consistent with the data at 2.4 GHz discussed above. Recall that the

11
channel was initially sampled at a much higher rate, down-converted to baseband, averaged to obtain an estimated representation of the signals received by each antenna, and then processed

according to the co-phasing algorithm. All the results presented here are from the processed data.

Figures 17 to 32 show the results from applying the co-phasing algorithm with the transmitter

located at the eight different test points in Building 24. Figures 17, 19, 21, 23, 25, 27, and 3 land

are the total power results versus the sample number. These results illustrate how the inclusion of

more array elements increases the total power. The 50 samples represent a total measurement

period of approximately 86 seconds.

The corresponding order in which array elements are added to the cumulative sum is depicted in

Figures 18, 20, 22, 24, 26, 28, 30, and 32. The RX antenna positions refer to a particular array

element as the ad hoc array is receiving in this case. The order number refers to the order in the

cumulative sum; that is, 1^' means the first element in the sum, 2" mean the second element in

the sum, etc. The elements are added into the cumulative sum from strongest to weakest

contributions, so the 1^' element is the strongest contributor, the 2"^^ element is the next strongest

contributor, and so on. Most of the test positions, which in this system represent transmitter

locations, exhibit little change in the ordering for the cumulative sum. Two factors that will

affect the order are people or objects moving through the array and comparable signal strengths

of the elements. In the Building 24 measurements, the change of ordering can largely be

attributed to the comparable signal strengths since there were no moving objects or people within

the array. For example. Figure 32 indicates some reordering between the 2" , 3"^

, and 4' array

elements, but the V^, 5*, and 6* elements remain the same for all 50 samples. This indicates that

the signal levels of the 2" , 3"^

, and 4' elements are nearly equal, and thus any slight change in

the measured signal will cause the order of contribution to the cumulative sum amongst these

three elements to change. The T', 5*, and 6* elements signals are sufficiently distinct that their

order of contribution does not change.

Figures 33 to 46 show the results for the seven fixed transmitter locations at the Denver site. Figures 33, 35, 37, 39, 41, 44, and 45 and are the total power results versus the sample number.

At the Denver site, 100 samples were collected over a period of approximately 173 seconds.

Note that at the Denver site, some collection positions include samples at the start when the

transmitter is not on. Figures 34, 36, 38, 40, 42, 43, and 46 indicate the order of the array

elements in the cumulative summation process. In general, a little more reordering appears as compared to the Building 24 case, but for each of the seven transmitter locations, the initially

selected receive element remains the same each sampling (ignoring the first few samples when

the transmitter is turned off). People were moving through the array area at the Denver site, and

this is the probable cause of changes in the element summation order.

At the Denver site, one experiment consisted of walking a handheld transmitter around the ad

hoc array area to capture some of the dynamic nature of the ad hoc array. Figures 47 and 48

show the co-phased cumulative sum and the order of the contributing receive elements,

respectively. 150 samples were collected over a period of approximately 255 seconds. In this case, the transmitter power was set to 30 dBm. Figure 47 exhibits greater than a 40 dB range.

The lowest total power, near sample 50, occurs when the transmitter was carried along the

12
sidewalk, well away from any of the array elements. Figure 48 shows significant changes in the

array element order, which is expected, because the transmitter location is changing.

4.3 Alternate Representation of Sampling Card Results

Averaging of several samples from the previous results for the sampling cards allows a more

direct comparison to the spectrum analyzer measurements. In Figures 49 and 50, the raw power

is plotted versus of the number of array receiving elements for Building 24 and the Denver site,

respectively. These values are computed by averaging the results from sample number 5 to 45 for

the Building 24 data, and sample numbers 15 to 85 for the Denver site (these data ranges where

chosen to avoid including any data collected during the off/on and on/off transitions of the

transmitter). The total power is generally greater than the spectrum analyzer data because of the

in-line amplifiers used with the sampling cards. (There are other loss factors not included with

the sampling card system, such as the power divider loss.)

For comparison purposes, the normalized cumulative power is plotted in Figures 5 1 and 52 for

Building 24 and the Denver site, respectively. Figures 51 and 52 compare to Figures 13 and 14,

respectively, though a true direct comparison is not possible, because the spectrum analyzer

results are for 2.4 GHz and the sampling card results are for 750 MHz. However, similar to the

spectrum analyzer results, for most of the test positions, the gain is between 2 to 6 dB when the

second element is included and the rate of gain increase beyond four elements drops

significantly. Test position 3 for the Denver site shows less than 2 dB of gain even at six array

elements because of the close proximity of the transmitter and the receiving array element

located at G. (Recall that this array element was not included in the spectrum analyzer data for

test position 3.) Test position 7 in Building 24 behaves nearly the same for both measurement

systems, which is likely due to the fact that the test position is elevated above the array plane.

4.4 Log-distance Path Loss Exponent Estimation by use of the Ad hoc Array

The data collected to demonstrate the array concept can also provide insight into the propagation

environment in the presence of the array. This analysis examines the path loss characteristics of

the environment, and in particular, estimates of the path loss exponent are provided. Reference

[16] contains relevant material on path loss calculations and typical results.

The following equation is used to estimate the path loss exponent, n,

P(J.)dBm = P, dBm - PL{dB) = P^dBm-\On\ogid./d,),

where / is the array element at a distance d. from the test location, d^^ is the distance between

the array element used as the reference and the test location, and P^^ is the measured power of the

reference array element. Since only one measurement was taken at each location, the t\pical

additional random variable is not included on the right side of (7), for example, see equation

(4.93) in [16]. Also, the path loss for the location do is accounted for in the measured power at that location, and that do is the location with the strongest contributing array element to the

13
among all possible test location. Thus, for the 2.4 GHz case in Building 24, the do is the distance

between array element E and test location 5; for the 2.4 GHz in Denver, do is the distance

between array element D and test location 7; and for the 750 MHz case in Denver, do is the

distance between array element G and test location 3. Note that a path loss exponent of « = 2 is assumed for the reference location, which corresponds to free space. The 750 MHz case for

Building 24 was not included, because a clear reference pair could not be established. This is because the array element location E was not included in the 750 MHz data collection, and all other possible combinations included objects or orientations in the path between the transmitter

and receiving element, so that there was no reasonable choice for Pq and the associated do. Note

that in order for (7) to be a valid estimate, di > do and P(di) < Po are required.

Figures 53 and 54 show the 2.4 GHz case for Building 24 and the Denver site, respectively. The

n values range from 1.1 to 5.7, which is consistent with results presented in [16]. Note that an n

value of less than 2 implies a waveguide affect. Further examination of Figure 53 reveals an

average of w = 3.6 for test location 7, which is due to the fact that the test location is not at the

same elevation as the array elements. The average of « = 3.1 at test location 8 is due to the

additional walls and obstructions between the array elements and the receive test antenna as compared to test locations 1 to 6.

Figure 54 indicates an average path loss exponent of between 1.9 and 2.4 at 2.4 GHz, which

more closely tracks the behavior of free space (n = 2) than the results from Building 24. This is expected, because the atrium area is a more open environment than the basement in Building 24.

The path loss exponent for the Denver building ranges from 1.1 to 3.2, as compared to a range of

1.3 to 5.7 for Building 24.

The 750 MHz path loss exponent estimations are shown in Figure 55. The average of 2.0 to 2.6

indicates a behavior slighdy worse than that of free space, but is consistent with the results for

2.4 GHz. Useful 750 MHz results were not available for Building 24, due to the lack of a good

reference position within the array topology relative to the test locations. The reference location

used for the 2.4 GHz case in Building 24 was not included in the 750 MHz measurements.

5. Measured Data versus Simulations

We tested the analytic expression in (5) against our measured data subject to several

approximations discussed below. As illustrated below, the simulations compare favorably with

the measured results. We compare the normalized results from the simulations and measured

data, because this removes the need to accurately predict absolute power levels, and because the

ad hoc array gain rather than the absolute power level is our quantity of interest. Thus the

following assumptions and parameter values are used:

1) A=\

2) / = constant

3) (Wc ^ 271 X frequency

4) Old = uniformly random over [0, In)

5) g)ir„ = adjusted to maximize r(t), in m discrete steps over [0, 2n); m = 16 for these

simulations

14
6) N= 6 orl, the number of ad hoc array elements

7) Li> 1 , (one corresponds to no multipath contribution)

8) an = Xldi, where di is the distance between the /''' array element and the test location

9) aik = {X/dkf^, where dk is uniformly random over [5, maximum of 2 x d^^ and n is uniformly random over [2.5; 6].

In these MISO simulations, the test and array element locations are those used in the actual

measurements, listed in Figures 9 and 10. These distances are the dominant factors in the

simulated array performance. The ot,/t values provide a very rough approximation of the

multipath contribution; more precise values can likely be extracted from other measured data. A

simulation run averaged 10,000 trials, where a trial consisted of selecting a random value for 9ki,

and aik over the intervals specified above, and then performing the co-phasing algorithm in the

simulation. Setting the time variable of ? as a constant means that the simulation results

correspond to a specific instant in time. Thus, the ajj variable becomes a constant phase offset. A more exact simulation would incorporate both the time and frequency dependence of aik,

although as previously noted, the channel can be considered time-invariant for a large fraction of

the time. Note that the results from the channel sampling measurement system indicated that

channel was generally not changing rapidly with respect to time, at least not enough to change

the order of the array elements in the co-phasing algorithm.

Simulation results based on the ad hoc array and test locations are shown in Figures 56 to 59.

Figures 56 and 57 are based on the Building 24 locations, while Figures 58 and 59 are based on

the Denver site. In Figures 56 and 58, no multipath contribution is included, which represents a

"free space" environment. Figures 57 and 59 include ten multipath components in addition to the

direct path. The most noticeable difference between the free space and multipath simulations is the spread of values for a given number of elements (2 or more). For example, the two element

case for Building 24 has a spread of approximately 4 dB for free space and a spread of

approximately 2 dB for the multipath case. Excluding test location 3, the Denver site exhibits a

similar decrease as multipath components are included. Both Building 24 and the Denver site show a decrease in the maximum for test location 4 (using six array elements) of approximately

1 dB in the multipath versus free space.

The simulation results correspond to the normalized ad hoc array measurements. For Building

24, Figures 13, and 51, compare to Figures 56 and 57, and for the Denver site, Figures 14 and 52

compare to Figures 58 and 59. In the Building 24 case, simulation results do not predict as wide

a spread of values, which is likely due to the lack of incorporating impacts of local obstacles on

the antenna gain pattern, and to inaccuracies in multipath amplitude and phase modeling.

Weighting due primarily to the distance would be more accurate when the antennas radiate in a true omnidirectional pattern associated with a point source. The rate of gain increase is similar

between the simulations and the measured data up to inclusion of the first four elements.

However, the measured results appear to reach a diminishing return point near the four-element

mark. This is due in part to the additional signals being near or below the noise floor oi the

measuring equipment, which is not accounted for in the simulation.

Simulations and measurements of the Denver site compare reasonably well, particularly w hen

the measurements are taken from the sampling card results (Figure 52). Note that the spectrum

15
analyzer measured results in Figure 14 do not include array element G for the case of test location 3, which dominates the other array elements in the simulations and the sampling card

measurements. As pointed out in the path loss exponent estimation above, the Denver site behaves much like a free-space environment, so the closeness of match with the simulations is expected.

6. Summary of Results and Conclusion

The primary focus of this effort was to measure the performance of an intra-volume ad hoc array

concept to improve public-safety communications. We also provided a mathematical framework,

different from the original optimization analysis, to assist communication engineers in applying

the proposed approach to wireless systems in a real-world setting. Simulations based on the

descriptive mathematics and some rough approximations indicate a reasonable prediction of the

ad hoc array behavior. The combination of the measurement results and the mathematical

framework serves as the basis from which to design a wireless ad hoc array system for public

safety.

Two different measurement systems tested the intra-volume ad hoc array concept in a transmit

mode (MISO), using mechanical phase shifters at 2.4 GHz, and in a receive mode (SIMO) with

channel sampling cards at 750 MHz. Measurements were carried out at two different building

sites, one in a large basement and the other in an atrium between two large office buildings.

Subject to differences in frequency dependencies of materials and the surrounding environment,

the array concept performed equivalently in transmit and receive modes, as expected.

Major measurement findings are: (1) the ad hoc array concept does provide gain performance

consistent with earlier predictions; (2) the gain typically ranged from 2 dB to 6 dB for two array

elements, and up to about 10 dB for four array elements; (3) the gain per element starts to reach

diminishing returns at around four elements, and (4) the coordinated use of only two elements can provide a significant gain improvement. The fourth point is important, because two wireless

nodes are generally easier to coordinate than three or more, and suggests that meaningful

improvement in public-safety communications can be achieved by use of two wireless nodes as a two-element array.

Further refinement of analysis in the presence of multipath propagation, and extension to modeling of modulated signals are two areas of future work. The natural next step in the process

is to develop wireless nodes that can implement the ad hoc array technique.

Disclaimer: Mention of any company names serves only for identification, and neither

constitutes nor implies endorsement of such a company or of its products by NIST.

This work was sponsored by the U.S. Department of Justice through the Public Safety

Communications Research Lab, Derek Orr, Program Manager. We thank members of the

16
technical staff of the Electromagnetics Division 818, who assembled the test equipment, and

Michael Francis and Perry Wilson for managerial support. We also thank Patrick Fine and his

staff for providing access to the 555 17'*^ Building in Denver, Colorado

17
7. References

[1] Statement of Requirements: Background on Public Safety Wireless Communications,

The SAFECOM Program, Department of Homeland Security, Vol. 1, March 10, 2004.

[2] M. Worrell and A. MacFarlane, Phoenix Fire Department Radio System Safety Project,

Phoenix Fire Dept. Final Report, Oct. 8, 2004,

http://www.ci.phoenix.az.us/FIRE/radioreport.pdf

[3] 9/1 1 Commission Report, National Commission on Terrorist Attacks Upon the United

States, 2004.

[4] Final Report for September 11, 2001 New York World Trade Center terrorist attack,

Wireless Emergency Response Team (WERT), Oct. 2001.

[5] C.L. HoUoway, G. Koepke, D. Camell, K.A. Remley, D.F. Wilhams, S. Schima, S. Canales, and D.T. Tamura, "Propagation and Detection of Radio Signals Before, During

and After the Implosion of a Thirteen Story Apartment Building," NIST Technical Note

1540, Boulder, CO, May 2005.

[6] C.L. Holloway, G. Koepke, D. Camell, K.A. Remley, and D.F. Williams, "Radio

Propagation Measurements During a Building Collapse: Applications for First

Responders," Proc. Intl. Symp. Advanced Radio Tech., Boulder, CO, March 2005, pp.

61-63.

[7] C.L. Holloway, G. Koepke, D. Camell, K.A. Remley, D.F. Williams, S. Schima, and

D.T. Tamura, "Propagation and Detection of Radio Signals Before, During and After the

Implosion of a Large Sports Stadium (Veterans' Stadium in Philadelphia)," NIST

Technical Note 1541, Boulder, CO, October 2005.

[8 ] K. A. Remley, G. Koepke, C.L. Holloway, C, Grosvenor, D. Camell, J. Ladbury, J. T. Johnk, D. Novotny, W. F. Young, G Hough, M. C. McKinley Y. Becquet, J Korsnes,

"Measurements to Support Broadband Modulated-Signal Radio Transmissions for the

Public-Safety Sector, " NIST Technical Note 1546, Boulder CO, April 2008.

[9] W. F. Young, E. F. Kuester, C.L. Holloway, "Optimized Arbitrary Wireless Device

Arrays for Emergency Response Communications," NIST Technical Note 1538, Boulder

CO, March 2005.

[10] W. F. Young, E. F. Kuester, C. L. Holloway, "Optimizing Arrays of Randomly Placed

Wireless Transmitters for Receivers Located Within the Array Volume," Antennas and

Propagation, IEEE Transactions on, vol.55, no. 3, pp. 698-706, March 2007.
[11] W. F. Young, E. F. Kuester, C. L. Holloway, "Measurements of Randomly Placed

Wireless Transmitters used as an Array for Receivers Located within the Array Volume

with Application to Emergency Responders," Antennas and Propagation, IEEE

Transactions on, vol.57, no.l, pp.24 1-247, March 2009.

[12] W. F. Young, "Optimizing Arrays of Randomly Placed Wireless Transmitters with

Receivers Located in the Array Volume," Dissertation, University of Colorado at Boulder, 2006.

[13] M. L Skolnik and D. D. King, "Self-phasing array antennas," IEEE Trans. Antennas

Propag., vol. 12, no. 2, pp. 142-149, Mar. 1964.

[14] L. D. DiDomenico and G. M. Rebeiz, "Digital communications using self-phased

arrays," IEEE Trans. Microw. Theory Tech., vol. 49, no. 4, pp. 677-684, Apr. 2004.

[15] G. D. Durgin, "Space-Time Wireless Channels," Pearson Education, Inc., Publishing as Prentice Hall, Inc, Upper Saddle River, NJ, USA, © 2003.

[16] T. S. Rappaport, "Wireless Communications: Principles and Practice, 2nd Edition,"

Chapter 4, Prentice Hall, Inc, Upper Saddle River, NJ, USA, © 2002.

[ 1 7] Cobra CompuScope 2 1 G8, GaGe Cobra^'^,

http://www.w^untronic.eu/index.php'?site=2&xid=65&subid=185&sub2id=186&pid=255

19
Appendix I: Experiment Setups and Locations

Transmitting

Monopoles

Transmitting

v.^ Monopoies

_ oSignal Generator/ Power Amp

Figure 1. Sketch of experimental setup where a signal generator, power divider, and series of

mechanical phase shifters provide power to transmitting monopoles mounted on individual

ground planes. The receiver setup consists of a monopole on a ground plane connect to a spectrum analyzer. 120 foot (=37 m) cables connect the transmitting monopoles to the phase

shifters. A 60 foot (=: 18 m) cable connects the receiving monopole to the spectrum analyzer.

This is the setup used for the 2.4 GHz measurements.

20
Figure 2. Pictures of equipment used for the 2.4 GHz experiments. Power is supplied to seven monopole antennas through a single signal generator and a power di\ ider. The

output of the power divider is either fed to an antenna or terminated in a 50 Q load. The

spectrum analyzer measures the received power at the single receive antenna, and the

phase shifters are used to adjust the relative phase between transmitting antennas.

21
Figure 3. Several of the 2.4 GHz antennas used in the 2.4 GHz experiments in the

basement of Building 24, both in the main room and the connected hallway.

22
Figure 4. Some of the 2.4 GHz antennas used in the 2.4 GHz experiments at the Denver location, both

inside and outside of the atrium area.

23
Receiving

Antenna

5 Transmitting

Antenna

Receiving

Antenna

Sampling

Cards

Figure 5. Sketch of experimental setup where a signal generator and mechanical phase shifters

provide power to the transmitting antenna mounted on a ground plane. The receiver setup consists of

six antennas on individual ground planes connected to computer with RF sampling cards. 120 foot

(=^37 m) cables connect the transmitting monopoles to the phase shifters. A 60 foot (=^ 18 m) cable

connects the receiving monopole to the spectrum analyzer. This is the setup used for the 750 MHz

measurements.

24
Figure 6. Measurement computer with sampling cards. Note that an in hne ampHtler

provides approximately 20 dB of gain before the waveform is sampled.

25
Figure 7. Some of the 750 MHz antenna array elements in the basement of Building 24.

26
Figure 8. Some of the antenna array elements used in the 750 MHz measurements carried

out in Denver.

27
Drawing not to scale

Array Element x(m) y(m) z(m) TestRX/TX x{m) y(m) z(m) A 7.4 -AH 0.0 1 29.9 -9.2 OO

B Q3 -9.3 OO 2 16.7 -ia2 OO

C LO -5.6 L5 3 B.1 -58 0.0 D 2a8 -L7 OO 4 5.0 -7.4 0.0 E 1L5 -Z2 OO 5 129 -LI ao

F ia3 Z5 OO 6 20.4 -L7 0.0 G 1Q5 9.9 0.0 7 3Q8 -5.7 L5

8 30 -10.5 0.0

Measurement

Equipment

28.8 m

Y7

Yi

V J

Figure 9. Location of transmitter/receivers in the basement of Building 24 on the NIST, Boulder

campus. The (x,y,z) values are given in the included table, with a non-zero value of z if the location is on one of the stair landings. The rectangular boxes contain testing locations and the ovals contain the ad

hoc array elements.

28
17'" street Xaxis

Vi

Reference x=0, y=0

ffs)

Y axis

Measurement

Equipment

^v^

V.

Drawing not to scale

Array Element x(m) y(m) Test RX/TX x(m) y{m) A 22.3 -6.4 1 -6.3 -6.3 B 12.2 -6.0 2 -6.0 -20.3 C -0.3 -7.2 3 -16.9 -20.3 D 7.4 -21.6 4 -3.2 -27.6

E 4.0 -27.5 5 -6.2 -33.8

F 2.6 -35.5 6 7.1 -33.1 G -18.7 -20.3 7 7.1 -22.3

<D

CO

Legend

Test locaton i ^1/

Array element / \i/. location

Figure 10. Location of transmitter/receivers at building in Denver, CO. The locations are given in (x,y) pairs. The rectangular boxes contain testing locations and the ovals contain

the ad hoc array elements.

29
Appendix II: Experimental Results

-a

0)o

-55

-60

-65

-70

-75

-80

^ ^--^ —

^^

^

3 12 3 4 5 6 7 8 Number of contributing elements

Figure 1 1 . Building 24 raw results from spectrum analyzer measurements. Seven

contributing transmitting antenna elements and eight receive antenna locations (RX#)

were used.

2 3 4 5 6

Number of contributing elements

Figure 12. Denver raw results from spectrum analyzer measurements. Seven contributing

transmitting antenna elements for five of the receive antenna locations, and six

transmitting antenna elements for the other two receive location were used.

30
2 3 4 5 6

Number of contributing elements

Figure 13. Building 24 cumulative sum of co-phased power. Power normalized by largest

contributing element.

2 3 4 5 6

Number of contributing elements

Figure 14. Denver cumulative sum of co-phased power, normalized by the largest

contributing element. Two cases, RX 3 and RX 6, include only six transmitting

elements.

31
s

-a

-70

-75

-80

-85

-90

-95

-100

-105

-110

-115

t^

' 1 1 1 1 ^

; ; Single element Single element ^ Noise measurement + Two co-phased + + 6

-

lI::a .""\

.' " > Sj^. + -

/'\

M.. +

••-,.''

1 1 1

v.../' w 1

1 2 3 4 5 6 7 8 9

Receive antenna position

Figure 15. Building 24 results for two elements: (a) received power for single element,

(b) increase in received power for two co-phased elements. Both elements were

located in the hallway, off the main room.

-/u

-75 ik

+

-80 '^..y '^ y

-

-85 + + ' 7^. -

£

-90 . / "\ 4-

CQ

/•"^, 1 -o -95 ; > ' _,.,

-100 " »

* #

k:>;'

\.,.*

-105

-110

1 1 c

'

- ; Single element ^ y Single element 4- Noise measurement —

|— Two co-phased -

1 1 1 1 1 1 1 1

Receive antenna position

Figure 16. Denver results for two elements: (a) received power for single element, (b)

increase in received power for two co-phased elements. Two of the weaker elements were selected, but not necessarily the weakest two.

32
15

-20

-25

?CQ

<L> O

B -35

e2

-30

-40

-45

iiL^^i|:i|:^5£-SMis^**^-W ^'

»•«'>•

10 20 30

Sample number

40 50

Figure 17. Building 24, absolute value of co-phased cumulative sum; TX

position 1.

a4

COC

cS 3

• 9 *

4- * * * -K- ^- + "i- -i- ^ «• -« HJf- !!« + 4- + 4" * -f * * t+^.^.-^.r*-^-i^+.u

1st 2nd ^ 3rd 4th 5th 6th

10 20 30

Sample number

40 50

Figure 18. Building 24, TX position 1. RX order in the co-phased summation.

33
15

-20

-25

-30

3

o

a.« -35 oH

-40

-45

T

»»-«.-* -«-»-«--«-«-*-«»-» ©»-»-« ««»«-&»-»--»-«(?* « a-a-gift- «-»--« »-s-..«-••»-

* '' *"i

-

-« 1 RX

2RXs

-^-3RXs

- Q 4RXs

5RXs —6RXs

-

r 1 1

-

10 20 30 40 50

Sample number

Figure 19. Building 24, absolute value of co-phased cumulative sum; TX position 2.

cac

u

n 3 X

Pi

5-^- -» ^ »* + + + ^ *** + + + + ^' ** lif ** + + + -"^ *** + ++ + ^^ **»*-»- + '++ •f' *+ + + + +

-e««««««9eft09«»9eo«««9««««a««sd»«««»«««««@««^«4e»»-

'l

• 1st 2nd ' 3rd 4th 5th ' 6th

1 1 1 1

2-uu

10 20 30

Sample number

40 50

Figure 20. Building 24, TX position 2. RX order in the co-phased

summation.

34
-IJ

-20

-25

oB -35

-40

-

- 1 RX

2RXs

-+~-3RXs

- 4RXs

5RXs —6RXs

-

-45

1 1 1 1

-

10 20 30

Sample number

40 50

Figure 21. Building 24, absolute value of co-phased cumulative sum; TX position 3.

a

d

S 3

9 » ^ ^ ®®»®»9*»#

. -I,- -+. -!|i. «• 4- •* *• •»- > -li- -* « » -•-* + + «-*»* -v - -+•+ 4- -»i<- •*»•+•+*** + •*+ -i '

• 1st 2nd » 3rd 4th 5th >< 6th

10 20 30

Sample number

40 50

Figure 22. Building 24, TX position 3. RX order in the co-phased summation.

35
-15

-20

-25

S

m

o

-30-

2 -35 oH

-40

-45

-jp. -+-.;.. .i_.i^.i..^...|.. J),...^_.i....i...w.^.j_t-..4_i)iv,^..i.j,...j5...i._.^^^

- -

-

» 1 RX

2RXs

- - 3 RXs

- 4RXs

5RXs —6RXs

-

1 1 1 1

-

10 20 30

Sample number

40 50

Figure 23. Building 24, absolute value of co-phased cumulative sum; TX position 4.

c

c 3

^ + + + + -f cj * * i;:; + 4^ +•*••» c ; + * -r •¥• -r * + + 4- + -«• -r * * O* + + ;:; -4- -r * ^^ +- -

1st 2nd - 3rd 4th 5th ^ 6th

10 20 30

Sample number

40 50

Figure 24. Building 24, TX position 4. RX order in the co-phased summation.

36
15

-20

-25

6

m

o

a. 3 -35 oH

-40

-45

>..^..^-a «•-««« fe-ft-a -»-*«-.« -<

10

«-«' «»»-:

» 1 RX

2RXs

^^ 3RXs

- 4RXs

5RXs —6RXs

20 30

Sample number

40 50

Figure 25. Building 24, absolute value of co-phased cumulative sum; TX position 5.

at

c

c

<u ^ 3

jji- ^- ^- » « * * *• 4' 'i- »: » •* ^- '!' -^^ •+ 4- -i' * *« « * » -^^^ * -i- 4- -i- -i- + -i- * '^-i- -^^ -t- + -i- -f ^- 4- *

!» » » « 9 8 S 4 e « « « « # 1

1st 2nd 3rd 4th 5th 6th

10 20 30

Sample number

40 50

Figure 26. Building 24, TX position 5. RX order in the co-phased summation.

37
15

-20

o

-25-

-30

« -35- oH

-40

-45

'"-+--^'-i--^ i*^-*^-i^H^—

«-»-«.-»-«--« '-»-#..,--*-«»©«-»' *"*«'-'

•^-s-*--*-®-^ -,...»._»..«-»«-<.'«-»«*«-'

» 1 RX

2RXs

-^-3RXs

- 4RXs

5RXs —^6RXs

10 20 30

Sample number

40 50

Figure 27. Building 24, absolute value of co-phased cumulative sum; TX position 6.

asC

G

S 3

c3 X

%i»«ai9@^&ftt»iaftafl& ee«aa«a«ie®*'

-r

-*- -^^ + -*- ^ -r -r -r -i^ -^^ -^ ^s- ^ «- -ije -*- -ir -i- -^e -^ -jje -*-«--»-*>-+. ^

1st 2nd 3rd 4th 5th 6th

10 20 30

Sample number

40 50

Figure 28. Building 24, TX position 6. RX order in the co-phased summation.

38
-15

-20

-25 ?pq3l-H o

Oh B -35 o

-30

-40

-45

't'T :^:$e«<»%9@9«-iS ,«.»«>*

* 1 RX

2RXs

" * 3 RXs

' 4 RXs

5RXs —6RXs

10 20 30

Sample number

40 50

Figure 29. Building 24, absolute value of co-phased cumulative sum; TX position 7.

^ 3 X

«^@@@a®9a«#s< «00»®<3««6ft'

--«-^-ie-^-V-)ir*-jie-Jie^-^-'je-f--^-*-**-«-* f** + + + *

1st 2nd + 3rd 4th 5th 6th

10 20 30

Sample number

40 50

Figure 30. Building 24, TX position 7. RX order in the co-phased summation.

39
-15

-20

-45

, IRX

2RXs

' 3RXs

-- 4RXs

5RXs —^6RXs

^+-+--<!.-»-*_^» *^-<i--^+-^

-o- «-<.»,. ,--»-^-s- *-.g- «^ * -® -a- ^-^

10 20 30

Sample number

40 50

Figure 31. Building 24, absolute value of co-phased cumulative sum; TX position 8.

§.4

^ 3

(S3

* * i- G * * a :*. +4- -4- •» * ai3 * * *- *'

««»*«*»«»«»&»«»*»

+ + * .* .» X * * + *. -.*. «- * >t * -:

««$S3^aa

« 1st 2nd • 3rd 4th 5th < 6th

10 20 30

Sample number

40 50

Figure 32. Building 24, TX position 8. RX order in the co-phased summation.

40
-15

-20

-25

B

OQ3;-! O

o.« -35 oH

-30

-40

-45

'« -^S*®®*© .*^ '^**®S®*'e,

»-. /^ _, -"^, 'V.A^ „ ^>s^^/^v^ '/'^/^A '/^

%«W*, ^ ^,'^©«»«« -^IRX

2RXs

~-*~ 3RXs

n 4RXs

5RXs

^6RXs

20 40 60

Sample number

80 100

Figure 33. Denver Building, absolute value of co-phased cumulative sum; TX position 1.

a4

(U S 3

c3

2L«

k-J(--lfrH-'^-^'i-f--^.^iiiM*^^ii-i-if-i^^^

i .0 (»t^9«<«^«eee@«si»0s«000e9«e«9@%®e«9«9««99i9«es(teae^««oeo44«ft»««iiie9«9e«««««4«ft««««e«s««»»e«««a*««««***i

>

1st 2nd * 3rd 4th 5th " 6th

20 40 60

Sample number

80 100

Figure 34. Denver Building., TX position 1. RX order in the co-phased summation.

41
-15

-20

-25

-30 o

ex 3 -35 o

-40

S(«^%S%*«®«r®«

-45

A X -

I I i ® » ^ IRX

2RXs

-^ 3RXs

4RXs

5RXs

-^6RXs

20 40 60

Sample number

80 100

Figure 35. Denver Building, absolute value of co-phased cumulative sum; TX position 2.

at

c 3 X

Pi 2

^4^:;;^i|:I5f-H-*-¥-*-H-*'^:i:N:!;i;:0!III^

Kff^'.li.^^.TlTi^K'ie^^.'^^ »v>^i^1HH^^'i^**¥**t"jVVy^^

1st 2nd * 3rd -' 4th 5th < 6th

20 40 60

Sample number

80 100

Figure 36. Denver Building, TX position 2. RX order in the co-phased summation.

42
15

-20

-25

m3

oa

B -35 oH

-30

-40

-45

«• <*\ «« » * »«*

^-1 RX

2RXs

* 3RXs

4RXs

5RXs —6RXs

20 40 60

Sample number

80 100

Figure 37. Denver Building, absolute value of co-phased cumulative sum; TX position 3.

a

0) X

Pi 2

iT^-^^-^i^ t- ^t^ ^ g y y jy y

*? i ^ ^ 'M'

'^-^^^^^^. m4*m**i*fv!-viinn'

1st 2nd ' 3rd 4th 5th - 6th

20 40 60

Sample number

80 100

Figure 38. Denver Building, TX position 3. RX order in the co-phased summation.

43
-20 - M^^^^^m^^^^^^^^^^^i^^^^^'^^^^^^^^^^^^^^

-25

S

CQ-a

- -

.•.,^^.^..*°»^«^^,^^^^^^'^.\».»,,^»»o%^,»,«a^«'^*,,^»=^«'^«/««^»»'»»»'^«»«^V»»«'*

fe-30 oB -35

~

* 1 RX

2RXs

e2 -^ 3RXs

-40

-

Q 4RXs

5RXs

-

" 6 RXs

-45

1 1 1

1

-

20 40 60

Sample number

80 100

Figure 39. Denver Building, absolute value of co-phased cumulative sum; TX position 4.

a 3

20

-Jfr]frif'if^-^-*.-^

-*--*-^>lrtf-Jfri>i-Sf-ih^ )jt K K-i^ih%-i^>ifri>rllfrJ^Jfrif-if-i!-)^-:ili^-^^

1st 2nd - 3rd 4th 5th 6th

40 60

Sample number

80 100

Figure 40. Denver Building, TX position 4. RX order in the co-phased summation.

44
-15

-20

-25 m3;-! O

:§ -35 oH

-30

-40

-45

"if^W^f^^^S^5:r^^;t<::, ,.:::v^^^j5j3h^^^^

V .'"^^^'/.^o-^Y**"'/^sar^es ,'»**»at!**8««»a!»<«»»«>et>«8««ae»*a«»»»«»esi,e»ese%"'* »,' jjiR«*««r •'

» 1 RX

2RXs

' 3 RXs

4RXs

- 5RXs -

-

1

-^6RXs

1 1

1

-

20 40 60

Sample number

80 100

Figure 41. Denver Building, absolute value of co-phased cumulative sum; TX position 5.

6 -nr*-

XXX a0£)Ae«>fii00»«)«)9e9»e<»0ae>«9«>0«)«0«99«a0e« a d>9#9t!«#««>»<s<£'«9se3e%«90»e<»'»e»»e«9«6eea««00««»»«e»«eB «»««««» i

c

c

c 3

CO

1st

20

2nd - 3rd 4th 5th

40 60

Sample number

80

6th

100

Figure 42. Denver Building, TX position 5. RX order in the co-phased summation.

45
-15

-20

-25

-30

en

o

a. B -35 oH

-40

-45

i»«*»«»»®*««®«ft«e»»a>8e*a&«»«*«.«sff»e«99»««»»*fi»e««»»»»»»»«»*»««effie©*a®**«*ae fi*« ^^««^g®^H8««««»**"

20

• 1 RX

2RXs

-^ 3RXs

a- 4 RXs

5RXs

^6RXs

40 60

Sample number

80 100

Figure 43. Denver Building, absolute value of co-phased cumulative sum; TX position 6.

-^i»x-«*-»*(»*=tofeaH|^H-+4^-*-*^^

99!»8»o»t»s»s«£«e«ss«£«»e»»«»e»te!ittses«4atta«««»st:aaesssdttee®«aa9e««a»tt»aeie^aa&«0««9*«««ie»ssa££atttt«aae9&«'«»«»i>

20 40 60

Sample number

1st 2nd + 3rd 4th 5th " 6th

80 100

Figure 44. Denver Building, TX position 6. RX order in the co-phased summation.

46
-15

-20

-25

-a

o

o,

;2 -35 oH

-40

-45

'•H^4+^4iM»)<i**'****++*+

»^»>i«»s»«a^»»,»»»«»»«M,«»«»e» »»«<!«*»«»<">»**"'**'"'»*'««*">««»,,.,.»««.,»»»..«,«„»,^«...^,^,„,^,

IRX

2RXs

* 3RXs

4RXs

5RXs

-^6RXs

20 40 60

Sample number

80 100

Figure 45. Denver Building, absolute value of co-phased cumulative sum; TX position 7.

40 60

Sample number

100

Figure 46. Denver Building, TX position 7. RX order in the co-phased summation.

47
-45 50 100

Sample number

150

Figure 47. Denver Building, absolute value of co-phased cumulative sum. Transmitter carried over path both inside and outside of the ad hoc array. Note that the vertical scale is increased

by 3 dB over previous Denver plots, e.g., compared to the vertical scale in Figure.

C

c

c 3

^IIa^.IIIi!;!:^!!X^s'^Bf' ''^^J^C?*

^:l01SH]5s<;-;fS>tnn^03O:^vx

:

r^^*j-" -H^»;«-»wp<^««»»4««ft 9 s®-^--»i- ijsajBas «B-^(- '^iijil-

:

iiihN'" *•«(*«»* ft r*-+w-®

«

l!*:**»^* -f -H=-3WSJ**::?»j -i^*^ ^ZiniOIMIIOIOiS!*^

e 3«s ®»»^>+»t'-^s»i'- »»a * H^; :^";'^n >*:" :>ocf ;t iT" X

1st 2nd 3rd 4th 5th 6th

50 100

Sample number

150

Figure 48. Denver Building, TX carried over path both inside and outside of the ad hoc

array. RX order in the co-phased summation.

48
-15

-20

-25

i -30

3i-i I

-35

-40

-45

-50

^

TXl ^^ TX2

H~ TX3

^e TX4

-^ TX5 ^^ TX6

-^ TXT ^

TX8

i^^'

^- —^— -^

^'

^-^

12 3 4 5 6 7

Number of contributing elements

Figure 49. Building 24 raw average cumulative power results based on 41 samples. TX

number refers to the transmitter position, which is the same as the RX position used in the

spectrum analyzer measurements.

2 3 4 5 Number of contributing elements

Figure 50. Denver raw average cumulative power results based on 71 samples. TX

number refers to the transmitter position, which is the same as the RX position used in

the spectrum analyzer measurements.

49
2 3 4 5 Number of contributing elements

Figure 51. Building 24 average cumulative power results based on 41 samples, normalized by

the largest contributing element. TX number refers to the transmitter position, which is the

same as the RX position used in the spectrum analyzer measurements.

12

10

m•T3

O

Oh •aN

O

2 3 4 5 Number of contributing elements

Figure 52. Denver average cumulative power results based on 71 samples, normalized by the

largest contributing element. TX number refers to the transmitter position, which is the same

as the RX position used in the spectrum analyzer measurements.

50
6.0

5.0

4.0

c 3.0

2.0

1.0

0.0

#

'>f $

'

A k

j^ 4 f

...i... i

J t- A

-X-

.'*: V1

H

..«.. X

••=_., » »«» » « • **"

# A

B 4 C

\ D

\ E • F

.4... 6

. _- Ave.

...„,,.. • Std. Dev. 12 3 4 5 6

test location

7 8

Figure 53. Building 24, 2.4 GHz path loss exponent for test locations 1 to 8, and array

element A through G.

6.0

5.0

4.0

c 3.0

2.0

1.0

0.0

• » « •

# A

B A C

\ D

r E

F

..... Q —»—Ave.

- >• Std. Dev.

2 3 4 5 test location

Figure 54. Denver building, 2.4 GHz path loss exponent for test locations 1 to 7. and

array element A through G.

51
6.0

5.0

4.0

c 3.0

2.0

1.0

0.0 i i T

2 3 4 5 test location

A

B

Ji. C

A E

F

,..;.., G

. _-Ave. —• Std. Dev.

Figure 55. Denver building, 750 MHz path loss exponent results. Note that array

element "D" is not included because only six channels are sampled with the

computer and sampling card system.

52
2 3 4 5 6

Number of contributing elements

Figure 56. Simulation results from for the Building 24 positions, with no

multipath contribution. Average of 10,000 trials.

2 3 4 5 6

Number of contributing elements

Figure 57. Simulation results for the Building 24 positions, with three multipath

components. Average of 10,000 trials.

53
2 3 4 5 6

Number of contributing elements

Figure 58. Simulation results from for the Denver site positions, with no multipath contribution. Average of 10,000 trials.

2 3 4 5 6

Number of contributing elements

Figure 59. Simulation results from for the Denver site positions, with

ten multipath components. Average of 10,000 trials.

54
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