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^ecknlccii vtote 280

A PEAK AC-DC VOLTAGE COMPARATOR

FOR USE IN A STANDARDS LABORATORY

L. A. MARZETTA

U. S. DEPARTMENT OF COMMERCE

NATIONAL BUREAU OF STANDARDS
THE NATIONAL BUREAU OF STANDARDS

The National Bureau of Standards is a principal focal point in the Federal Government for assur- ing maximum application of the physical and engineering sciences to the advancement of technology

in industry and commerce. Its responsibilities include development and maintenance of the national

standards of measurement, and the provisions of means for making measurements consistent with

those standards; determination of physical constants and properties of materials; development of methods for testing materials, mechanisms, and structures, and making such tests as may be necessary, particularly for government agencies; cooperation in the establishment of standard practices

for incorporation in codes and specifications; advisory service to government agencies on scientific and technical problems; invention and development of devices to serve special needs of the Govern- ment; assistance to industry, business, and consumers in the development and acceptance of commercial standards and simplified trade practice recommendations; administration of programs in cooperation with United States business groups and standards organizations for the development

of international standards of practice; and maintenance of a clearinghouse for the collection and

dissemination of scientific, technical, and engineering information. The scope of the Bureau's

activities is suggested in the following listing of its three Institutes and their organizational units. Institute for Basic Standards. Applied Mathematics. Electricity. Metrology. Mechanics. Heat. Atomic Physics. Physical Chemistry. Laboratory Astrophysics.* Radiation Physics. Radio Standards

Laboratory:* Radio Standards Physics; Radio Standards Engineering. 0£Bice of Standard Reference

Data. Institute for Materials Research. Analytical Chemistry. Polymers. Metallurgy. Inorganic Materials. Reactor Radiations. Cryogenics.* Materials Evaluation Laboratory. Office of Standard Refer- ence Materials. Institute for Applied Technology. Building Research. Information Technology. Performance

Test Development. Electronic Instrumentation. Textile and Apparel Technology Center. Technical

Analysis. Office of Weights and Measures. Office of Engineering Standards. Office of Invention and

Innovation. Office of Technical Resources. Clearinghouse for Federal Scientific and Technical

Information.**

* Located at Boulder, Colorado, 80301. Located at 5285 Port Royal Road, Springfield, Virginia, 22171.
NATIONAL BUREAU OF STANDARDS

Technical Note 280

ISSUED JANUARY 17, 1966

A PEAK AC-DC VOLTAGE COMPARATOR

FOR USE IN A STANDARDS LABORATORY

L. A. Marzetta

Electronic Instrumentation Section

Measurement Engineering Division

National Bureau of Standards

NBS Technical Notes are designed to supplement the Bureau's regular publications program. They provide a means for making available scientific data that are of transient or limited interest. Technical Notes may be

listed or referred to in the open literature.

For sale by the Superintendent of Documents, Government Printing Office

Washington, D.C., 20402 - Price 25 cents
Contents

Page

1, Introduotlon 1

2, Study of Peak Yoltage Detecting Circuits 4

2.1 Use of Diode Rectifier 4.

2.2 Smith and Clothier Comparator 5

2.3 Resistance Summing Comparator 6

3* Description of NBS Peak ac-dc Transfer Standard 7

3.1 Relay Synchronizer S

3.2 Mplifier 10

3.3 Power Supplies 11

4. Test Results 12

Table 1 15

Table 2 16

Table 3 17

Table 4 Ig

List of Figures 19

II
A Peak ac-dc Voltage Comparator

For Use in a Standards Laboratory

Louis A, Marzetta

An increased availability of high-precision a-c and d-c power sources has allowed routine electrical

measurements to be made in the field with a precision

previously possible only in the standards laboratory.

Ac-dc comparisons are being made on the basis of the

peak, rms, and average value of the sine wave source. The peak ac-dc voltage comparator described in this

report was developed for use in the standards laboratory to help meet the increasing demand for the certification of new field instruments. The described instrument is capable of establishing the ac-dc error of

other peak comparators at frequencies from 50 to 2^00 Hz

with an imprecision of less than 15 ppm.

Key words: Voltage, alternating, direct, calibration,

comparator, standard, peak and transfer,

1. Introduction

A brief look into the past history of electrical measurement could

be helpful in understanding the evolution of instrumentation for use in

a standards laboratory. The structure for the derivation of electrical

units gained a firm foundation with the efforts of Gauss, Weber, and

later. Maxwell, who in l86l headed a program for correlating electrical

and mechanical units. In the early use of electricity, such as telegraphy,

quantitative measurements involving resistance and direct current received

the greatest attention. Later in the 1880 's the use of electricity took

a heavy turn in the direction of power and lighting. Alternating current

electrical energy generation and distribution lagged that of direct current by only about one year. Thus the needs for ac-dc transfer techniques and instr'jjnents were established. For the next 60 years or so d-c

measurements held a very wide advantage in precision over a-c measurements. Great improvements had come about in resistance measuring bridges.

Around the turn of the century Weston perfected an electrochemical cell

vhich offered the standards laboratory a d-c reference voltage source Ith accurate level predictability and long term constancy.

1.
In 1958, F. Hermach (l) writing on the state of precise a-c measurement stated, "The accuracy of a-c measurement is often limited by the

stability of the source rather than the instrument." Until recent years

the source of a-c power at audio frequencies in standards laboratories

was the rotary alternator. Despite the handicaps with such sources of

sine wave voltages, laboratories around the world had achieved high precision rms a-c measurements. At NBS, the work of F. K. Harris (2) and

F. B. Silsbee (3) in the 1930 's resulted in the perfection of electrodynamic instruments such that ac-dc difference measurements could be made

with a precision of 0.01^ at power frequencies and about 0.1^ at frequencies

close to 3 kHz. The rms value of measurement was a natural selection

since the majority of physical effects in experiments and other usage are related to the transformation of electrical energy to heat dissipated in

a current carrying resistor or to mechanical forces developed between

current carrying conductors or electrically charged plates. Since harmonic contamination affects rms measurement very little, moderate wave form distortion was not very troublesome at that time. Total harmonic

distortion of a few percent could be tolerated when making accurate determinations. The precision of measurement with electrodynamic instruments

had reached a leveling-off stage, and by nature their frequency response

was severely limited.

The next stage in the progress of precise a-c measurement came in

the field of electro-thermal ac-dc transfer instruments, especially the

contributions by F. Hermach of NBS in the 1940 's and 50 's (l), (4), (5).

His development of the differential thermocouple transfer standard opened

the door to measurement precision of better than 0.01^ throughout the

audio range of frequencies and extended precise a-c determinations well

into the radio frequency spectrum. This work came at a time when it was

(1) F. L. Hermach, "AC-DC Transfer Instruments for Current and Voltage

Measurements", IRE Transactions on Instrumentation, pp. 235-240,

Dec. 1958.

(2) F. K. Harris, "A Suppressed Zero Electrodynamic Voltmeter", J. Res.

N.B.S., Vol. 3, PP. 445-457, 1929.

(3) F. B. Silsbee, "Composite Coil Electrodynamic Instruments", J, Res.

N.B.S., Vol. 8, pp. 217-264, 1932.

(4) F. L. Hermach, E. S. Williams, "A Wide-Range Volt-Ampere Converter

for Ciirrent and Voltage Measurements", AIEE Transactions, pt. 1

(Co^-nmunications and Electronics), Vol, 78, Sept. 1959, pp. 384-388.

(5) J. E. Griffin and F. L. Hermach, "A Differential Thermocouple Voltmeter", AIEE Transactions, (Communication and Electronics), Vol. 81,

No. 63, pp. 339, November 1962,

2.
needed not only to enrich the capabilities of the many standards laboratories but also to meet the urgent requirements for precise a-c measurements brought on by rapid advances in guidance equipment associated with

the defense buildup.

In 195/^ W. Smith and W. Clothier (6) of NPL, Australia, reported a measurement study that although it did not attract much attention at that

time may now be classed as a pioneering work of significant contribution

to a-c measurement. Although the USA, as well as laboratories in other

countries, had for years concentrated on electrodynamic transfer instruments, the Australians and the British had placed their faith in an

electrostatic voltmeter transfer standard. They believed that by overcoming the surface contamination problems of the capacitor plates, the

electrostatic voltmeter could offer ac-dc transfer precision of better

than 0.01^, (lOO ppm). What is most revealing in the paper is that

Smith and Clothier proposed to use a peak-to-peak ac-dc voltage comparator (with an electronic amplifier and oscilloscope) as the primary means

of a-c to d-c transfer determination, and then to use the values so obtained to establish the ac-dc error difference of the electrostatic

Instrument, (More details will be given later on this procedure.) In

order to make use of peak-to-peak measurement, the authors had to give

serious attention to the control of the amplitude stability and waveform

purity of the alternating potential source. With a well designed electronic oscillator, feedback amplifier, and harmonic filter, their a-c

source had an amplitude stability of 0,001^ per minute and a total harmonic

content of 0.001^ at a single frequency of 50 Hz. The overall performance

of the system resulted in a measurement precision of 0.001^ to 0,

A number of years elapsed before electronic techniques and devices

were used with ac-dc standards. In 1962 P. Richman presented one of

several papers that appeared on this subject (7) (8). Sparked by demands

of the military, Richman and others ushered in the era of "Electronic

Standards" with wide range, audio frequency, power standard generators

having total harmonic distortion of less than 0.01^, with a similar

figure for week to week amplitude stability, and d-c supplies with 0.005^

stability for long periods of operation. Other instruments included peak

ac-dc voltage comparators and an average ac-dc transfer standard. The

(6) W. E. Smith and W. K. Clothier, "Determination of the D.C./A.C.

Transfer Error of an Electrostatic Voltmeter", The Proceedings

of the Institution of Electrical Engineers, Part 2 (Power Engineering),

No. 83, pp. 465-469, Oct. 1954.

(7) P. Richman, "A New Peak-to-DC Comparator for Audio Frequencies",

IRE Transactions on Instrumentation, Vol. 1-11, Nos. 3 and 4,

p. 115, Dec. 1962.

(8) P. Richman, "A New Absolute AC Voltage Standard", Paper 13,

Session 26, IEEE Record of Convention held March, 1963,

3.
work described in this report was initiated to help meet the increasing

demand for certification of new instruments,

2. A Study of Peak Voltage Detecting Circuits

If an instrument is to be developed for the purpose of comparing the

peak value of a sine wave with an equivalent d-c value, the instrujnent

must have an electronic circuit that performs some type of signal transformation at the time of the sine wave peak. The following is a brief

review of a number of designs that have been considered, including a few

statements of their relative merits or shortcomings.

2.1 Use of a Diode Rectifier as a Peak Voltage Detector

For a rectifier with almost ideal characteristics, the d-c output

voltage of Figure 1 will be close to the peak value of the input sine

function for the polarity in the forward conduction direction. Near

ideal characteristics in a rectifier imply a forward rectifier resistance

that is orders of magnitude lower in value than the load resistance R , and a reverse direction resistance orders higher than R . In addition

o to this, the R C time constant must be many times greater than the period

of the input waveform, and the rectifier must have no contact potential.

Consideration of the output time constant, R C, is necessary in

order to establish the desired degree of accuracy in converting peak a-c

to d-c. The droop of the d-c output from one sine peak to the next

is an exponential decay of the form e o . If the load time constant

is very much greater than the input period, T, the droop is almost linear

and the d-c value for half-wave rectification will be in error by one-half

of the droop percentage. For a few parts per million accuracy, the load

time constant must be several thousand times the input function period;

a situation that leaves the overall system with a very slow response time.

Slow response coupled with ever present amplitude instability at the source and extraneous noise will lead to serious errors. One compromise

would be to reduce the R C time constant and accept a computable error.

As it turns out, the problem stated is academic since available

diodes are far from ideal. A test of the high quality solid state diodes

reveals a forward resistance value of 10 ohmS for a voltage of 3 milli6 volts; in fact, the forward resistance is still 10 ohms at 0.3 volts for

a silicon diode. In addition the same diode suffers from back conduction.

At about 20 millivolts potential the reverse resistance is only twice the

forward resistance. Thermionic diodes have better resistance characteristics but possess a high unstable contact potential of about 0.8 volt.

4.
Before giving up on the peak rectifier circuit, two modified arrangements should be mentioned. If the diode rectifier in Figure 1 is replaced by a synchronous mechanical switch or by a synchronized ring diode

gate, the dilemma of a long charging time constant and short system response time can be resolved.' With a synchronous switch, the long output

time constant can be kept in the interest of low droop at the output; yet

the circuit can retain the advantage of cycle to cycle peak charge restoration. It might be said that the system has both fast response and

long term peak memory storage. The price involved for this innovation is

a switch that can open and close for the briefest period exactly at the

peak of the sine wave; a duration in the vicinity of one-half degree of

angle. Some form of impedance translating circuit would have to be used

in order to transfer the d-c potential at the high resistance R C network

to an output with a low resistance for subsequent measurement of the d-c.

Possibly a highly stabilized electrometer amplifier could be used.

2.2 The Smith and Clothier Time Selection

Peak ac-dc Voltage Comparator

Smith and Clothier (6) were probably first in recognizing the potentiality for precision peak measurement with the development of a nearly

pure sine wave a-c source. With such a source made available, they

designed a peak-to-peak ac-dc voltage transfer device to be used as a standard at an a-c frequency of 50 Hz. The end purpose was to use the

peak comparator as a means for determining the ac-dc error of an electrostatic voltmeter which was itself used as a working transfer standard.

The principle of their peak-to-peak comparator is shown in Figure 2.

A circuit was developed by means of which a high speed relay can be synchronized with the sine wave signal, V, , closing its contacts briefly at

each alternation peak. Two stable d-c sources, Vb and Vbp, serve the

purpose of shifting the average position of the sine wave, V , so that

each peak when alternately examined by the synchronous relay will have

its level positioned near reference ground. The values of Vb and Vbp

are adjusted by watching the switching pattern on an oscilloscope. It

consists of a sequential positive and negative peak pulse train. Proper

adjustment of Vb^ and Vbp is evident by both polarity peaks precisely

touching a reference line on the scope representing ground potential.

With the completion of this peak-to-peak to d-c voltage transfer adjustment, the electrostatic voltmeter is alternately connected from the a-c source .V-, :.;to V, which represents a known portion of the sum of Vb.

and Vbp. Any difference in the deflection of the electrostatic voltmeter

for the two comparisons represents the ac-dc transfer error of the voltmeter. The error magnitude can be determined by readjustment of the V,

5.
calibrated voltage divider, R , at the output of Vb^ and Vbp for an ac-dc

difference of zero at the voltmeter. The difference between the adjusted

fraction of voltage at the voltage divider and 1/2V2 is the ac-dc transfer error of the electrostatic voltmeter. The 2 V2 term comes from having

initially set each of the two dc voltages to the peak value of the sine wave. Test results at 50 Hz indicated an agreement between the peak-topeak comparator and the rms electrostatic voltmeter with an average precision of better than 0.003^, and at no time did the data show a transfer

error of greater than 0,01^,

While Smith and Clothier had demonstrated the potentiality for precision a-G measurement with instrumentation of a basic electronic nature,

a number of years went by before their ideas were expanded and development of instruments for commercial distribution became a reality.

2.3 A Resistance Summing Peak ac-dc Voltage Comparator

Recognizing the increasing need for precise a-c measurement in both

the military field and in the standards laboratory, industry has responded

within the last five years with a remarkable number and variety of precision instruments. These include highly stable d-c power supplies, very

low distortion a-c supplies, and a good many measuring devices. Among

the latter is a peak ac-dc voltage comparator developed by P, Richman (7),

for use over a frequency range of 50 Hz to 20 kHz, The basic circuit is

shown in Figure 3. In the interest of versatility and wide frequency

response the design avoids the use of a time-selection approach for ex- amining the peak of each sine wave. Instead, it uses an ac-dc resistancesumming network with amplitude limiters which allow the viewing of a narrow slice of amplitude instead of time. As with the Smith-Clothier

system, an oscilloscope serves as the null indicator.

Considering the circuit of Figure 3 in more detail, the selected

polarity peak of the sine wave is shifted, with the aid of an identical

pair of summing resistors, to a common reference by a d-c voltage of the

opposite polarity. A back-to-back pair of silicon diodes connected at

the summing junction clips off the signal excursion at approximately the

0,7 volt level in both polarity directions. The relay in the circuit

switches at some uniform rate between the clipped sine wave peaks at the

summing junction and the common reference line, which may be ground.

After further amplification the waveform is again amplitude limited before

application to an oscilloscope. The waveform at the scope face consists

of a group of sine wave peaks appearing as sharp spikes in sequence with

the zero reference line, A null balance is achieved by varying the d-c source in such a manner as to bring the sharp peaks to the same reference

level. In this way a peak ac-dc transfer has been made. By reversing

the d-c source polarity the other a-c peak can be compared, and averaging

the two d-c values will remove the effect of even-order harmonic distortion

that may exist in the a-c source. This author suggests a variation in

scope presentation. According to usual instructions the scope sweep is

6.
synchronized with the relay at the line frequency, 60 Hz, producing the

sequential pattern of spikes and reference line described previously.

However, if the scope is synchronized from the a-c source, the sweep rate

can be increased to the point where one expanded peak of the sine wave

tip appears superimposed on the zero reference line. The advantage of

this presentation is that there appears to be less confusion about setting

the sine peak and the reference line into correspondence with ever present

noise on both functions. Of added importance is the fact that the expanded function allows a detailed study of the waveform which may be influenced by noise and other disturbances. When used with near ideal a-c

and d-c sources, the described peak comparator can give consistent results with a precision of 0.005^ over a wide frequency range,

3, Description of the NBS Peak ac-dc Transfer Standard

With an a-c source having known values of harmonic contamination, it

is possible to calibrate a peak a-o to d-c instrument using existing rms thermal transfer standards. However, it would be more realistic and

time saving to perform such a calibration using a laboratory peak transfer

standard.

The basic idea in mind in the development of a new peak comparator

was the search for a circuit that would offer a simple, direct comparison of the peak of a sine wave and a d-c source. It was expected that

the desired simplicity would place some restrictions on operational

flexibility, but the ultimate need was for a laboratory standard and not

a field instrument.

In the design of the instrument the use of precision summing resistors for the a-c and d-o sources has been avoided; rather, the a-c and d-c

voltages are time sequentially compared to a common reference. Eliminating

the summing resistors removed a possible source of error that would have

been present with unknown variations in their relative values. Also, with

a resistive network there is the possibility of an ac-dc error due to reactive components both within the resistors and external to them.

The system uses a pair of synchronized relays to slice out a time

section of the sine wave at its peak in the manner of the Smith and

Clothier scheme. These relays have time-overlapping closure, and are synchronized from binary frequency dividers and phase shifters. They

extend the range of measurement to several thousand Hz, The electronic

amplifier was designed for simplicity. All possible storage elements,

such as capacitors, have been deleted to give fast recovery and to minimize the frequency sensitivity of the circuit. External feedback loops

with their usual reactive components have been avoided in order to achieve

a flat frequency response with a near natural upper frequency roll-off

characteristic

,

Figure 4 is a block diagram showing the fundamental operation of the

NBS instrument and a graph of the overlap operation of the relays is

7,
included in Figure 5. The overlap operation simulates a single-pole

double-throw switching unit which samples the a-c for less than 100 microseconds, a requirement which no single relay could meet in synchronized

operation. Selection of relay design was made on the basis of several

requirements unique to this circuit. The chosen relays would have to be

of the high speed type, that is reliable operation in excess of 100 cycles

per second. Relays capable of high speed operation would maintain better

synchronism and also require fewer frequency divider stages when operating

with a-c input frequencies of several thousand hertz. Relay designs with

minimum contact bounce should be given preferential consideration. Long

operating life is also important; the devices should function without

degraded operation for millions of operating cycles. In the interest of

stability at high speed operation, a relay excitation coil with low winding

inductance and resistance would be easier to fit into a circuit design.

The type of relay that best met the requirements was the commercially

available reed relay with mercury-wetted contacts. They can be obtained

for speeds of up to 200 operations per second, with less than 50 ohm

windings, 50 milliwatt operating power, and an operating life of more than 10 cycles. In addition, non-bridging contacts could be specified

in order to avoid a common electrical path in the switching circuit

between the a-c and d-c sources.

3.1 Relay Synchronizer

Figure 6 is a schematic diagram of the transistor circuit necessary

to synchronize the relay operation with respect to the a-c source. The

circuit can best be understood by following the signal flow from input

to output with a brief explanation of the function of each electronic

stage.

The input of the synchronizer system is the same a-c voltage that

is to be compared with the d-c voltage. Silicon diodes are used to clip

both polarities of the sine wave leaving a 0.7 volt square-wave-like

signal at the input of an emitter follower, QlOl, A Schmitt trigger

circuit made up of Q102 and 0,103 and designed with triggering levels

close to zero reference is able to generate a large square wave output

with rise and fall excursions that are close to the zero crossings of

the input sine wave. QIO^- is a phase-splitter stage with two outputs

one in phase at the emitter and the other output at the collector having

a 180 degree phase relationship with the input. In a peak to d-c com- parator it is desirable to examine both polarity peaks of the sine wave. In the system being described, a relay under the control of the d-c source polarity is used to select the proper phase output of QIO4 for

subsequent synchronizing purposes.

The instrument was designed to operate over a frequency range from

50 to 2400 Hz, Since electro-mechanical relays cannot function at the

higher speeds, six binary stages have been incorporated in the circuit

for the purpose of frequency dividing the a-c input, A seven-position

switch allows the selection from 1:1 to 64.rl frequency division; the
selected position is chosen for a relay operating speed of about 50

operations per second. For example^ in the case of an input a-c signal

of 2400 Hz, one would select the 64 ;1 division position. In this mode

every 64th peak of the a-c source would be examined by the narrow time

gate. No information is lost by this procedure; but the oscilloscope

must have sufficient trace intensity to display an expanded pattern with

a duty cycle that may be less than 0.1^. Referring to Figure 6^ the

differentiated output of the binary selector switch is directed to emitter

follower, Q117, which in turn operates two single-shot multivibrator

stages, Q119, QX20, Q121, and Q122, Each multivibrator stage is used to

generate a controllable time delay. At this point the system branches

off into two independent channels for operating relay A and relay B,

Since the channels are identical, only one need be discussed. The time

delay in single shot multivibrator^ 0,119 and Q120 is under the manual

control of a front panel 10 turn precision potentiometer with digital

dial for logging purposes. Both time delay multivibrators are initiated

at the same time but with independent turn off times set by the precision

potentiometers. The delay circuit serves two purposes — one function is

to control operation of the relay in order to center the generated time

gate exactly at the sine peak, and secondly, to stagger the two delay

times in order to form a sampling time gate of the desired width as shown

in Figure 5, Logging of the digital dial settings helps in setting up

the instrument for each new change of input frequency. The differentiated

output of Q120 and the output of the reset signal handling stage Q118 are

used to trigger the binary stage Q123 and QI24 which is used to generate

the waveform for subsequent excitation of the relay coil, Q127 is the

output stage for driving the power amplifier.

The relay drive circuit is shown in Figure 7. Q201 is a power amplifier for the signal output of Q,127 in the previous schematic diagram.

Figure 6. Coupling from the last stage, Q201, to the relay coil is made

through a capacitor, 0201, for the purpose of shifting the normally negative going voltage at the collector into a potential that swings both

positive and negative with respect to common. This is done in order to

improve the on-off operation of the single-side stable relays. The series

resistor between capacitor and relay coil was selected on the basis of

reliable operation with minimum current in the interest of high speed

synchronization. As shown in Figure 5, the normally "OFF" side of the

relay switching was chosen as the common "ON" position for the overlap

operation. In this mode less contact noise was observed. The combined

operation of the two relays acts as a single-pole double-throw switch

that makes a synchronized closure on the a-c side for a time as short as 50 microseconds and a much longer closure time on the d-c side. Relay

instability or "jitter" has been less than 10 microseconds in practice.

Since the instrument sequentially samples the a-c peak and the d-c

source, the polarity of the peak must correspond to the d-c polarity.

Some arrangement had to be designed into the system for translating the

chosen 10-volt signal level down to the zero-state common reference. An

isolated, stable, well filtered 10-volt supply is located between the

9.
relay output and the amplifier input. To simplify operating procedure,

a polarity sensing circuit has been included which consists of stage

Q203 and a pair of common variety relays Rb3 and Rb^., The transistor

senses the d-c source polarity and operates the relays in an appropriate

manner. One relay establishes the polarity of the lO-volt series supply,

while the other selects the proper phase polarity at QIO4 in Figure 6 in

such a manner that the delay circuits will position the time gate on the

desired polarity peak.

3,2 Amplifier

The purpose of the amplifier is to provide the proper transfer circuit between the output of signal-gating relays and the oscilloscope

input. A first requirement in considering a choice of design is the

degree of amplification itself. For oscilloscopes having a vertical

deflection sensitivity between 0.01 and 0.1 volts per cm, the instrument

amplifier should have an amplification figure of from 10 to 100, Loading

effects of the amplifier input on the a-c and d-c sources had 'to be considered, since the degree of measuring precision is to be close to 10 ppm

(0.001^). Keeping the loading error below this figure implies an amplifier input impedance of at least 100,000 times the impedances of the two

sources. Although modern precision a-c and d-c supplies may have source

impedances of much less than 0.1 ohm, the amplifier was designed with an

input impedance greater than 100,000 ohms.

Another aspect of the amplifier characteristic is the question of

bandwidth. Since the a-c input frequency extends from 50 to 24.OO Hz,

at first glance one might be tempted to assign a modest frequency response requirement to the amplifier. In order to avoid an amplitude

correction factor greater than 5 ppms, the bandwidth of the circuit must

be such that the upper corner frequency is at least 300 times the highest

signal frequency involved. A bandwidth in excess of 1 MHz was assigned

as a requirement in the amplifier under discussion.

It can be assumed that low frequency roll-off is as pertinent to

signal amplitude integrity as the high frequency case. This problem was circumvented by designing the amplifier to have d-c response, A very

low frequency corner, by the use of large coupling time constants, would

satisfy the bandpass requirements but would result in a bouncy pattern

at the oscilloscope, due to the base line shift with varying signal amplitude. While on the subject of d-c coupling some provision is required

for adjusting the average level of the amplifier output, i,e,, the amplifier should have a low d-c offset between input and output levels. This

is a circuit bias design problem.

Biasing of the first stage in the amplifier takes on more significance

with the present application than would be usual for normal operations.

The first stage should be biased in a manner that would leave the input

terminal close to zero potential with respect to common reference. Ordinary solid-state stages may have quiescent input potentials of a volt or

10,
more due to biasing. Since the signal selecting relays are of the breakbefore-make type, there are two periods in each cycle of operation when

the amplifier input is not connected to either signal source; if the

quiescent input potential were other than near zero, the effect would be

manifested as abrupt voltage transients at the amplifier output.

Figure S is the schematic diagram of the amplifier designed for use in the NBS transfer instrument. It represents a simple circuit that

meets all of the stated requirements. The price paid for the simplicity

has been a compromise of amplification stability and quiescent d-c output

level stability, neither of which affects the instrument accuracy.

The first transistor, 0,301, presents an impedance of more than

100,000 ohms to the signal source while providing an ideal low impedance

output for driving the following amplifying stage. Q301 is biased in a manner that allows signal excursions in both polarity directions, and

also has a quiescent input potential of less than 35 millivolts. The

second transistor, Q302, is the voltage amplifying stage with an amplification figure of between 10 and 20 from d-c to about 1,5 MHz, A pair of

transistors, 0,303 and Q304, are used to provide an amplifier output impedance of less than 100 ohms; they also offer a symmetrical dynamic

signal excursion range with minimum loading to the amplifying stage. In

order to avoid the possibility of overloading the plug-in amplifier at

the oscilloscope (many of which are now transistorized), a pair of germanium diodes have been placed at the amplifier output to limit signal

excursions to less than one-half a volt.

3,3 Power Supplies

The word "ground" is conspicuously absent in the text of this report;

in its place the word "common" has been substituted. In the various

schematic diagrams the ordinarily used ground symbol has been replaced

by a new symbol resembling a large "G", for common. Instead of using the

instrument chassis as a common reference or ground, as is usually done,

this instrument has a common reference that is electrically isolated

from the chassis. The only connection to the chassis is the grounding

wire in the 3-terminal power line plug. Two purposes are served - one is

for safety from electrical shock, and the other is to allow the chassis

and cabinet to act as a grounded electrical shield. Electrically isolating

the circuitry from the chassis helps considerably to reduce the problem

of interfering signal currents in ground loops. This problem is sometimes

called "common mode effect". The problem has become more acute with the

advent of the 3 wire power plug. If the instrument has one side of its

signal lead grounded (as is the usual case), an interconnection of instruments may introduce unwanted ground currents into the signal circuit.

Ironically, some manufacturers recommend plugging the 3 wire power plug

into an adapter with a 2 prong output, in this way frustrating the original

intent of the 3 wire plug. The indiscriminate use of ordinary coaxial

connectors for low-level signals has led to much difficulty from ground

loops. The braided shield of the cable carries ground currents as well

11.
as signal currents because its connectors make conductive contact with

the chassis.

Additional paths for interference in many instruments are found

within the power supplies^ especially at the power transformer, Capacitive coupling from primary to secondary windings can result in noise

being transferred from power line into electronic circuitry, despite the

rectifier filters. To minimize this problem the three power supplies

in the described instrument have been selected with very low capacity

between windings and from windings to case. Figure 9 shows additional

circuits that are required to adapt the power supplies to the needs of

the system.

4, Test Results

The following tests were made in the Electrical Instrument Section

of the Electricity Division. 'A thermal ac-dc voltage comparator with an ac-dc transfer error of 10 ppm or less was made available and was used

as the standard in the following intercomparisons.

Figures 10 A and 10 B show how the instruments were connected. Both

the a-c and d-c supplies are high precision instruments with amplitude

stability of better than 25 ppm for extended periods of operation; their

noise content and the harmonic content of the a-c unit were less than

50 ppm»

Table 1 lists the results of an intercomparison between the InIBS Peak

ac-dc Voltage Comparator and a resistance summing peak comparator. For

this group of tests the a-c supply was set for a fixed value of 10 volts

peak amplitude. By means of oscilloscope observation of each comparator's

output, the amplitude level of the d-c supply was adjusted to imtch the

peak a-c voltage. The 6 dial reading at the d-c supply was recorded for

each determination. Results in the table show the difference in d-c dial

readings in parts per million (ppm) between sequential comparisons of the

two peak comparators for each frequency, A positive sign indicates a higher d-c reading by the NBS comparator over the other for the same a-c

value. Results shown cover 4- different a-c frequencies. Each determination is an average of the readings for both polarity peaks of the a-c sine wave. This procedure nearly eliminates the peak error effect due to evenorder harmonics present in the a-c voltage.

Note that all the difference readings in Table 1 are in the same direction. In all cases the resistance summing peak comparator assigned

a lower d-c value to the a-c peak than did the NBS peak comparator. The

discrepancy was established by additional intercomparlsons Involving the

thermal transfer standard, and will be described in the following paragraphs. One further remark on the results listed in Table 1 is the fact

that no one test value departed by more than 20 ppm for each frequency

from the averaged values; most were less than 10 ppm.

12.
r

The results of the previous tests demonstrated the precision or re- peatability of the peak oomparators but no conclusions could be drawn as to their absolute ac-dc t^ransfer accuracy. By comparing each peak com- parator with the thermal ac-dc standard instrument, the ac-dc error of

the peak comparators could be found. In addition, by taking the difference

of the ac-dc errors of the peak comparators as judged by the thermal

standard and comparing it with the difference readings described in Table

1, a cross check of all measuring instruments will have been completed.

The second and third group of tests concerned the comparison of the

peak comparators and the thermal transfer standard. Since these tests

involve measurement of both the peak value and rms value of the a-c voltage, the need for waveform purity is apparent. During these tests

the a-c output was kept at a constant 10 volts, rms. AVzT ratio transformer was included in the test circuit in order to normalize the 10 volt

values between the peak reading instrument and the rms thermal instrument.

The peak a-c value was compared with the voltage of a calibrated d-c

supply, using the peak-reading instrument in conjunction with an oscilloscope. The 6 dial reading of the d-c supply was recorded. The same d-c

and a-c values were next compared by means of thermal transfer standard,

and any difference between the two voltages was read at a calibrated

galvanometer at the output of the thermal instrument. Recorded results

in Tables 2 and 3 show the difference in ppm between the a-c and d-c

values as interpreted by the NBS Thermal Standard for a group of four

frequencies. Table 2 represents results with the NBS peak comparator

while Table 3 is for the resistance summing peak comparator. A positive

sign indicates a higher a-c value than the d-c one as measured by the

thermal instrument. As was the case with the first group of tests, the

individual values in most cases departed less than 20 ppm from the average value.

On the basis of the results shown in Table 2, the N3S Peak Comparator

has an ac-dc transfer accuracy of better than 0.0015^ (15 ppm) for all

the operating frequencies except 50 Hz. The maximum error occurred at a frequency of 50 Hz, the particular frequency at which the a-c supply and

output transformer has the poorest waveform purity.

Since the intended purpose for the NBS Peak Comparator is the direct

measurement of the ac-dc error in other peak comparators, an evaluation

of the accuracy of the data in Table 1 is in order. Now treating the

NBS Peak Comparator as a standard, the results in Table 1 show the magnitude and direction of the ac-dc error for the resistance summing peak

comparator. From Tables 2 and 3 the ac-dc difference between the two

peak comparators can be judged as follows:

= (fQo resistance summing comp.) i^Q.--) NBS peak Comp.

)

E Error diff. ( ^

. )-r-^ ) (ac-dc error ) (ac-dc error )

13.
The ability of the MBS Peak Comparator to establish the ac-dc error in another peak comparator is shown in Table <4. In this Table the ac-dc error of the resistance summing comparator judged by the NBS Peak Comparator is compared with the error established by the NBS Thermal Standard.

The agreement of the ac-dc error is better than 10 ppm at all four frequencies.

The author wishes to thank Dr. Don Flach of the Electrical Instruments Section, Many hours of exacting tests were undertaken in the

Standards Laboratories of that section in order to evaluate the described

instrument. The high order of precision of the comparison tests results

from Mr. Flach 's careful contribution to the program. The author also

thanks Mr. F. L. Hermach for his continued interest in the work and for

having originally proposed the problem.

u.
TABLE 1

F^ECiUENCY -- Hz

50 ^00 1000 2^00 RUN

d-c +16 + 6 + 27 + 21 1

Difference between +26 +16 +50 +41 2

peak comparators +21 +16 + 51 + 51 3

PPM +11 +16 +36 +46 4

§^) Average

PFM +19 +U +a +40

NBS peak comparator vs res summing peak comparator

istance

15.
TABLE 2

ac-dc

Difference

PPM

(^P ) Average

PPM

FREQ.IffiNCT - Hz

50 400 1000 2400 RUN

+ 26 -2 + 6 +2 1

+46 +3 -12 +16 2

+ 34 -16 -18 -23 3

+37 -11 -18 -4 4

+ 36 -7 -11 -2

NBS peak oomparator vs NBS thermal

standard^

16.
TABLE 3

ac-dc

Difference

PPM

FREQUENCT - Hz

50 400 1000 2400 RUN

+ 48 + 9 + 28 + 58 1

+ 51 +12 + 37 + 37 2

+46 + 16 + 22 +25 3

+35 +19 +15 +36 ^

j5^ ) Average

PPM +46 +14 +23 +39

Resistance summing peak comparator

vs NBS thermal standard.

17.
TABLE 4

C from TABIE 1

PPM

ITO:QUENGy - Hz

50 400 1000 2400

+19 +14 +41 +40

.S3 --^2 ^^°"

TABLES 2 and 3

PPM +10 +21 +34 +41

-Si- C53-&) +9 -7 +7 -1

Agreement of established error in

the resistance summing peak com- parator by the two methods.

IS,
LIST OF FIGURES

1. Half-Wave Diode Peak Rectifier.

2. Smith and Clothier Peak to Peak ac-dc Transfer

Standard

3. Resistance Summing Peak ac-dc Comparator

4-. Block Diagram of NBS Peak ac-dc Comparator

5, Relay Operation of NBS Comparator

6, Relay Synchronizing Circuit

7, Relay Driver

B, Amplifier

9. Power Supplies

10. Test Circuits

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