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.c'' NBS TECHNICAL NOTE 1099

REFERENCE

NBS

PUBLICATIONS

U.S. DEPARTMENT OF COMMERCE / National Bureau of Standards

Electromagnetic Compatibility and

Interference Metrology

M.T. Ma

M. Kanda
Tm he National Bureau of Standards' was established by an act of Congress on March 3, 1901. The ^^ Bureau's overall goal is to strengthen and advance the nation's science and technology and facilitate their effective application for public benefit. To this end, the Bureau conducts research and provides: (1) a

basis for the nation's physical measurement system, (2) scientific and technological services for industry and

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The Bureau's technical work is performed by the National Measurement Laboratory, the National

Engineering Laboratory, the Institute for Computer Sciences and Technology, and the Institute for Materials Science and Engineering

.

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Technology

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Conducts research and provides measurements, data, standards, reference

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'Headquarter!, and Laboratories at Gaiihersburg, MD, unless otherwise noted; mailing address Gaithersburg, MD 20899. -Some divisions within the center are located at Boulder, CO 80303. 'LtKated at Boulder, CO, with some elements at Gaithersburg. MD.
Electromagnetic Compatibility and

Interference Metrology

M.T. Ma

M. Kanda

Electromagnetic Fields Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Bureau of Standards

Boulder, Colorado 80303

.^^"^ '' ^0 /^ V# \

U.S. DEPARTMENT OF COMMERCE, Malcolm Baldrige, Secretary

NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Director

Issued July 1986
National Bureau of Standards Technical Note 1099

Natl. Bur. Stand. (U.S.), Tech Note 1099, 178 pages (July 1986)

CODEN:NBTNAE

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1986

For sale by the Superintendent of Documents, U.S. Governnnent Printing Office, Washington, DC 20402
CONTENTS

Page

Abstract 1

Chapter 1 . INTRODUCTION 1 REFERENCES H

Chapter 2. OPEN SITE 5 2.1 Site Attenuation Calculations 5 2.2 Mutual Impedance Calculations 7 2.3 Measurement Technique and Results 9 2.H Error Estimate 10

REFERENCES 10

Chapter 3. TRANSVERSE ELECTROMAGNETIC ( TEM ) CELLS 17

3 . 1 Impedance 18

3.2 Higher-Order Modes 18

3.3 Field Distribution 19

3.U Performing Radiated Susceptibility Tests 19

3.5 EUT Scattering in a TEM Cell 21 3.6 Performing Radiated Emission Tests 23

3.7 Examples of Some TEM Cell Applications 25

REFERENCES 26

Chapter 4. SHIELDING EFFECTIVENESS OF MATERIALS H^

it. 1 Theory—Small Aperture Coupling 44

4.2 Effects of the Receiving Antenna Orientation 46

4.3 Effects of Incident Field Distribution 47

4.4 Material Factors 48

4.5 Measurement Techniques 49

REFERENCES 53

Chapter 5. ANECHOIC CHAMBER MEASUREMENTS 65

5.1 Measurement of the Net Power to Source Antennas 65

5.2 Near-Zone Gain of Open-Ended Rectangular Waveguide 66

5.3 Near-Zone Gain Calculations of Rectangular Pyramidal Horns 67

5.4 Reflections from Anechoic Chamber Walls 69

REFERENCES 69

Chapter 6. OPEN-FIELD MEASUREMENTS 74

6.1 Standards for Field Strength Meters, 10 kHz to 10 GHz 74

6.2 Magnetic Field Strength Standards for Loops, 10 kHz to 50 MHz 74

6.3 Electric Field Strength Standards for Dipoles,

25 to 1 ,000 MHz [1 ],[7] 75

6.4 Electric Field Strength Standards for Vertical Monopoles,

30 kHz to 300 MHz (Also Standard-Field Method) 77

REFERENCES 79

Chapter 7. EM PROBE DEVELOPMENTS 90

7 . 1 Earl ier Models 90

7.2 Recent Models 91 7.3 Future Models 92

7.4 Diode Model 93

7.5 Application of Electro-Optic Techniques for Measuring EM Fields 94

REFERENCES 98

Chapter 8. REVERBERATING CHAMBERS Ill

8.1 Some Design Considerations Ill 8.2 Recent Measurement Results 113

REFERENCES 115

1X1
Chapter 9. CONTEMPORARY EMC TOPICS 129

9. a MEASUREMENTS IN, AND OF, COMPLICATED ELECTROMAGNETIC ENVIRONMENTS. .. 1 29

9.a.l An Electric-Field Meter (EFM) in a Complex Environment 129

9. a. 2 Characterization of Complex Environments 132

REFERENCES 135

9.b OUT-OF BAND EMC PROBLEMS U5

9.b.1 Reflector Antennas 145

9.b. 2 Coax- to-Wave guide Adapter 1 ^47 REFERENCES 1 U8

9.C CONDUCTED EMC PROBLEMS 155

9. C.I Characterizing CEMI Sources 155

9.C.2 Reducing CEMI Coupling by Grounding Computer Facilities 157

9.C.3 Measuring CEMI Emission and Susceptibility Levels 158

REFERENCES 1 60
ELECTROMAGNETIC COMPATIBILITY AND INTERFERENCE METROLOGY

edited by

M. T. Ma and M. Kanda

Electromagnetic Fields Division

National Bureau of Standards

Boulder, CO 80303

The material included in this report is intended for a short course

on electromagnetic compatibility/interference (EMC/EMI) metrology to be

offered jointly by the staff of the Fields Characterization Group

(723.03) and the Interference Characterization Group (723.0^1) of the

Electromagnetic Fields Division (723). The purpose of this short course

is to present a review of some of the radiated EMC/EMI measurement

methods, to which the National Bureau of Standards (NBS) at Boulder,

Colorado, has made significant contributions during the past two decades.

The technical foundation for these methods, and interpretations of the

measured results are emphasized, as well as strengths and limitations.

The entire course is presented in nine chapters with the

introductory part given as Chapter 1. The particular measurement topics

to be covered are: i) open sites (Chapters 2 and 5), ii) transverse

electromagnetic cells (Chapter 3), iii) techniques for measuring the

electromagnetic shielding of materials (Chapter H) , iv) anechoic

chambers (Chapter 5), and v) reverberating chambers (Chapter 8). In

addition, since small probe antennas play an important role in some of

the EMC/EMI measurements covered herein, a separate chapter on various

probe systems developed at NBS is given in Chapter 7. Selected

contemporary EMI topics such as the characterization and measurement of

a complex EM environment, interferences in the form of out-of-band

receptions to an antenna, and some conducted EMI problems are also

briefly discussed (Chapter 9).

Key words: anechoic chamber; complex electromagnetic environment;

conducted electromagnetic interference (CEMI); electromagnetic

compatibility (EMC); measurement methodology; open sites; out-of-band

responses; probes; radiated electromagnetic interference (EMI);

reverberating chamber; shielding of materials; transverse

electromagnetic (TEM) cell.

Chapter 1. INTRODUCTION

Electromagnetic compatibility/interference (EMC/EMI) may not be household words yet, but

their effects are widely apparent because of the prolific use of electronics and digital

circuits in a variety of products (television sets, personal computers, microwave ovens,

automobiles, etc.). Over the years, the undesirable effects of the electromagnetic

environment have been described by several terms such as radio frequency interference (RFI),

radio noise, electromagnetic pollution, EMC and EMI. The IEEE Dictionary defines EMC and EMI

as follows [1 ]

EMC: The capability of electronic equipment or systems to be operated in the intended

operational electromagnetic environment at designed levels of efficiency.

EMI: Impairment of a wanted electromagnetic signal by an electromagnetic disturbance.

One of the first recorded cases of EMI was documented in 1927 by the Federal Aviation

Agency [2], They observed that an aircraft altimeter was giving faulty readings and the

problem was caused by interference generated by the ignition system of the aircraft. The

first EMC military standard was published in 19'15, entitled "Interference Measurements, Radio,

Methods of 150 kc to 20 Mc (for components and complete assemblies)" [3].

Some electronic equipment such as transmitters are designed to supply radio-frequency

(RF) power at selected frequencies to antennas, from which the signal is radiated

intentionally. Signals which are radiated at other than the intended frequencies are
undesirable and called spurious emissions, a form of EMI. Other electronic equipment such as computers are designed to function as nonradiators. Ideally, a computer performs its tasks

and all signals generated within the system are contained and not radiated. However, in

practice, some of these internal signals may be radiated as EMI.

The process of establishing the electromagnetic compatibility of electronic equipment

usually requires two steps. First, measurements are made to determine if any undesired

signals being radiated from the equipment (radiated EMI) and/or appearing on the power lines,

control lines, or data lines of the equipment (conducted EMI) exceed limits set forth by the

using agency. This kind of measurement is commonly referred to as emission testing. The

second step is to expose the electronic equipment to selected levels of electromagnetic (EM)

fields at various frequencies to determine if the equipment can perform satisfactorily in its

intended operational environment. This process is referred to as susceptibility or immunity

testing.

The limits or requirements set forth for EMC testing are usually established by

regulatory agencies such as:

1) Federal Communications Commission (FCC), USA,

2) Department of Defense (DoD), EMC Military Standards,

3) Interdepartmental Special Committee on Radio Interference (IRAC), USA,

4) National Center for Devices and Radiological Health (NCDRH), USA,

5) Verboard Deutscher Elektrotechniker (VDE), West Germany,

6) Other foreign agencies.

In addition to the specifications by these regulatory agencies, there are a number of

voluntary EMI standards available for use. These standards are usually published by

professional organizations that utilize the expertise of many who donate their time and effort

in the preparation of documents. A few examples of these organizations are:

1) American National Standards (ANS) Committee C63 on Radio-Electrical Coordination, and

Committee C95 on RF Radiation Levels for Personal Safety,

2) Institute of Electrical and Electronics Engineers, Inc. (IEEE), EMC Society Standards

Committee,

3) Society of Automotive Engineers (SAE),

H) Electronic Industries Association (EIA) GM6 Committee on EMC,

5) Radio Technical Commission for Aeronautics (RCTA)

,

5) International Special Committee on Radio Interference (CISPR), and

7) International Standards Organization (ISO).

There are numerous measurement methods available for making radiated EMC/EMI tests

depending on the following considerations:

1) size of the test equipment,

2) frequency range,

3) test limits,

M) types of fields to be measured (electric or magnetic),

5) polarization of the field, and

6) electrical characteristics of the test signal (frequency or time domain).

Because each method has limitations of one kind or another, no one method is ideal for

all tests. Certain regulatory standards such as MIL-STD ^62 [iJ] or FCC part 15 [5] do specify

a particular method.

The purpose of this short course is to review some of the radiated EMC/EMI measurement

methods, in which the National Bureau of Standards (NBS) at Boulder, Colorado has been

involved in the past two decades [6]. The technical foundation for these methods, and the

interpretation of measured results are emphasized, as well as strengths and limitations. The

entire course is presented in nine chapters with the introductory part as Chapter 1. The

specific measurement methods to be covered are:

1

)

open sites (Chapters 2 and 6)

,

2) transverse electromagnetic (TEM) cells (Chapter 3),

3) techniques for measuring electromagnetic shielding of materials (Chapter 4),

4) anechoic chambers (Chapter 5), and

5) reverberating chambers (Chapter 8).
In addition, since small probe antennas play an important role in some of the EMC/EMI

measurement methods covered in this short course, a separate chapter on various probe systems

developed at NBS is included (Chapter 7). Furthermore, selected contemporary topics such as the characterization and measurement of a complex EM environment, interferences in the form of

out-of-band receptions by an antenna in modern communication systems, and some of the

conducted EMI problems are also briefly discussed (Chapter 9). Necessary laboratory

demonstrations at the NBS site (if the course is conducted in Boulder) showing how to make the

required measurements are also considered an integral part of the course.

Chapter 2, covers open site facilities. An open site consists of a large ground plane

which provides a well-defined EM environment for making measurements. Such measurement

facilities have been in use for a long time, and are still required for complying with testing

regulations set by many agencies and committees. They can be employed to perform both

radiated emission and susceptibility tests. The fundamentals such as the effects due to

mutual impedances between the transmitting and receiving antennas involved in the measurement,

the degree of impedance mismatches between the antenna terminal and a receiver, insertion

loss, various definitions for site attenuation, some practical measurement procedures, and

comparisons between actual measurement results and theoretical predictions will be discussed.

This chapter was contributed by R. G. FitzGerrell (723.0^1) and M. Kanda (723-03).

A related chapter on use of an open site for performing standard calibration services and

other measurements from 10 kHz to 10 GHz is given in Chapter 6. Both the electric-field and

magnetic-field standards, the definition of antenna factor, a simple model for theoretical

computation of the field strength produced by those commonly used antennas, and measurement

arrangements will be given. This chapter was prepared by E. B. Larsen and M. Kanda of 723.03.

Chapter 3 contains a discussion of transverse electromagnetic (TEM) cells. A TEM cell is an expanded section of transmission line used to create an isolated testing environment for

low- frequency applications. The cell's characteristic impedance, cut-off frequencies, higherorder modes, detailed field distribution inside the cell when it is used for susceptibility

testing, resonant frequencies, measurement considerations for both emission and

susceptibility, and interpretations of measurement results will be covered. This chapter was prepared by M. T. Ma, M. L. Crawford, and P. F. Wilson, all of 723.04.

The topics of measuring the electromagnetic shielding effectiveness of materials,

comparisons of measurement results by various techniques, and interpretation of results based

on theoretical models will be discussed in Chapter k. This chapter was contributed by P. F. Wilson, J. W. Adams and M. T, Ma, all of 723.04.

Chapter 5 deals with anechoic chamber measurements. The prime objective of using an anechoic chamber is to simulate a free-space condition over a finite test volume inside the

chamber for testing and calibration purposes. Methods of determining net power supplied to a source antenna and necessary near-field corrections in transmitting antenna gains will be

presented. This chapter was contributed by M. Kanda, R. D. Orr , and W. J. Anson of 723.03.

A summary of the development of various probe antennas at NBS for accurate measurements

of EM fields is given in Chapter 7. Earlier and recent models, their special features,

considerations for future models, a suggested mathematical modeling for diodes included in the

measurement system, and application of electro-optic techniques will be discussed in this

chapter. This chapter was contributed by M. Kanda and K. D. Masterson of 723.03.

Chapter 8 presents a relatively new measurement tool, namely, the reverberating chamber.

The main purpose is to generate a statistically average field throughout a test zone inside

the chamber for various high-frequency applications. The field strength level so generated is

usually very high. The basic principle involved in this facility, theoretical considerations

for designing a reverberating chamber, measurement specifics, the chamber quality factor,

characterization of the field environment created inside the chamber, and interpretations of

measurement results will be covered. The chapter was contributed by M. L. Crawford and M. T. Ma of 723.04.

Chapter 9 summarizes three selected contemporary EMI topics: the characterization and

measurement of a complex electromagnetic environment contributed by J . P. Randa and M. Kanda

of 723.03; out-of-band EMI problems contributed by D. A. Hill of 723.04; and some conducted

EMI problems contributed by B. A. Bell of the Electronic Instrumentation and Metrology Group

(722.02) of the Electrosystems Division (722) in Gaithersburg, MD.
REFERENCES

[1] Jay, F. ed. IEEE Standard Dictionary of Electrical and Electronics Terms, ANS/IEEE Std

100-1977, 2nd ed., New York, NY: Wiley and Sons; 1977.

[2] Donaldson, E. E. ; Free, W. R. ; Robertson, D. W. ; Woody, J. A. Field measurements made in

an enclosure. Proc. IEEE, Vol. 66(M): 464-^472; 1978 April.

[3] Jan-I-225, Joint Army-Navy Specification. Interference measurements, radio methods of,

150 kilocycles to 20 Megacycles (for components and complete assemblies). 19^5 June 14.

[4] MIL-STD-462. Electromagnetic interference characteristics, measurement of. 1967 July 31.

[5] FCC methods of measurement of radio emission from computing devices FCC Rules, pt. 15,

Subpt. J, Appendix A (July 1981) and FCC 83-352 order on general docket 80-284, Appendix

B. 1983 July.

[6] Ma, M. T; Kanda, M; Crawford, M. L; Larsen, E. B. A review of electromagnetic

compatibility/interference measurement methodologies. Proc. IEEE, Vol. 73(3): 388-411;

1985 March.
Chapter 2. OPEN SITE

Traditionally, open-site measurements provide a straightforward approach to evaluating

the EMI characteristics of the electronic equipment under test. An ideal site consists of

obstruction-free plane ground and the hemisphere above it, both infinite in extent. The

quality of measurements made using natural earth as the site depends on the ground constants.

To obtain more reliable measurement results, a metal ground plane or screen of a reasonable

size is normally recommended. The size used at NBS is a 30m x 60m wire mesh ground with an approximate lower frequency limit in the neighborhood of 20 MHz. One important factor to be

determined in application of this facility is the site attenuation or insertion loss between a source and a receiver. The site attenuation is herein defined as the ratio of the input power

to the transmitting antenna (or the voltage at the source signal generator) to the power at

the load impedance connected to the receiving antenna (or the voltage appearing across the

load). Thus, this definition includes mismatch losses but not balun or cable losses. As will

be seen later, once the site attenuation is determined accurately, the characteristics of an unknown emission source can then be evaluated by the measurement results from a standard

receiving antenna. The information so obtained is, of course, important for assessing the

electromagnetic environment created by this unknown emitter.

To facilitate the evaluation of this factor, two polarization-matched antennas (one for

transmitting and one for receiving) are set up above a ground plane as shown in Fig. 2.1. The

following parameters must also be specified: horizontal separation distance between the two

antennas; antenna type and polarization; and impedances of the transmitting and receiving

systems connected to the antenna terminals. Sometimes, the relative insertion loss defined as

the power indicated by the receiver at an initial receiving antenna height compared with that

at different heights, is evaluated instead. Under this situation, site attenuation is then

defined as the mini mum relative insertion loss measured between the terminals of two

polarization-matched antennas located on the test site when the receiving antenna is moved

vertically over a specified height range while the transmitting antenna height remains

unchanged [1]. Site attenuation is then determined as a function of frequency and horizontal

separation distance between the antennas for a given transmitting antenna height above the

ground. In this chapter, all the measurements are made from the NBS wire mesh ground screen

and results are presented in accordance with this latter definition.

Site attenuation measurement data obtained from a given site can be used to compare with

those from a standard or reference site to assess the site quality. Deviation between the

measured site attenuation and the theoretical result based on an ideal site then serves as an indication of site imperfection.

2.1 Site Attenuation Calculations

Considering an antenna system shown in Fig. 2.1 from the circuit viewpoint and assuming

that the reciprocity principle applies, we have

where

and

^ = ^1 ^11 *" ^1 ^11' ^ h h2 *" h ^12'

-^2 ^ = ^1 ^12 ^^^

^1 ^1-2 ^ ^2 ^22 ^« h ^22'

Z = mutual impedance between #r and #s antennas

rs

(r' and s' indicate images of #r and #s antennas),

V = input voltage for #1 (transmitting) antenna,

I. = input current for #1 antenna,

I„ = output current for #2 (receiving) antenna,

Z = impedance of the receiver attached to the feedpoint of antenna #2,

a = ground reflection coefficient.

Application of the relation Z = Z to (2.1) yields

Z = input impedance of #1 antenna = V /I

(2.1)
and

Z^^ * a Z^^, - (Z^2 " « ^1'2^^^^^22 ^ " ^22' ^ ^r^ '

Z = input impedance of #2 antenna

= Z22 - a Z22, - (Z^2 *" 2l.2)'/(^n ^ « ^11' ^^s^ '

(2.2)

(2.3)

where

Z = impedance of the signal generator, connected to the feedpoint of antenna #1

.

The power supplied to the transmitting antenna (#1) is then given by

P^ = 1 |lj2 Re(Z.^) , (2.H)

the voltage induced at the output terminal of the receiving antenna (#2) by

^2 = ^/^12 ^«Z^.2) ^« ^2^22- = ^1 ^m ' ^^'^^

where

^m = ^12 ^ "^1-2 ^« ^V^l^ ^22' • (2.6)

The maximum possible power accepted by a matched receiver is

In accordance with the site attenuation measurement method described in the I. E.G.

document [2], we have

2 S = I. E.G. (power) site attenuation = P /P

= ^ Re(Z.^) Re(Z.2) llj^/jv^l^ . (2.8)

Note that the above relation is true only under the ideal (conjugate match) condition. In

practice, this condition is certainly not realistic, because the input impedances, Z and

Z , are varying as a function of horizontal separation distance and antenna heights while the

receiver load may be a constant.

From the definition of the site attenuation measurement method described in the FGG

document [3], the site attenuation (voltage) is given by

1 I^S* ^ill-Is *^i2l • l^ll FGG 2 Z^ . IV^I ' ^ ^^'

R 2 Where Z is the characteristic impedance of a coaxial cable connected to the transmitting

antenna, Z is that connected to the receiving antenna, and V„ is given in (2.5).

n 2

If the impedances of the transmitting and receiving antennas are matched at both

terminals of the coaxial cables, and both antennas are identical and have the same height

above the ground plane, Z_ = Z., = Z.„ = Z^, = real (resistive only), the (power) site

o 11 12 K attenuation defined by I. E.G. becomes equal to the square of the (voltage) site attenuation of

F.G.G., i.e..
4ec = 4cc • (2.10)

In practice, the coaxial cable and antenna terminal are, most likely, not matched. Then

mismatch loss has to be incorporated into the calculation. Because of possible mismatch, the

net power out of the transmitting antenna, P' , is given by

Z . - Z P; = Pinn ^^ /^ ^ / ] . (2.11)

t inc Z.^ + Zg

where P. represents the power supplied to the transmitting antenna.

The actual input current, I' , then becomes

I| = [P^ / Re(Z.^)]^^2 , (2.12)

and the open-circuit voltage induced at the feedpoint of the receiving antenna is

V- = |I| Z^I , (2.13)

where Z is given in (2.6) m ^

The actual power received by the receiver is

P; = Re(Zj^) [V^ /\Z^ * Z.^l]^ . (2.14)

Since Z is a function of antenna height, frequency, and ground constants, its value

changes accordingly. For a fixed Z , frequency, and ground, we record the maximum value of P' when the receiving antenna height is changed. The site attenuation (based on power

consideration) is then defined as

sl^. = 10 log [P. /(P-) ] dB . (2.15) NBS inc r max

2.2 Mutual Impedance Calculations

Since the site attenuations defined in either (2.8), (2.9) or (2.15) involve the mutual

impedance, we now show how to calculate the mutual impedance between dipole antennas. For the

two monopoles above a perfectly conducting plane shown in Fig. 2.2, which may be considered to

represent two dipole antennas in free space, the mutual impedance between antenna #1 and

antenna #2 is defined as

^21 " ' ^2/^1^°^ • (2.16)

where 1.(0) is the base current in antenna #1, and the open-circuit voltage V produced at

antenna #2 by 1.(0) is given by

^21 = yo) ^/^z21 V^) "^ ' (2.17)

with Ip(z) as the current distribution along antenna #2, E as the parallel component of the

incident field along antenna #2, and h„ as the length of antenna #2.

The current distribution on antenna #2 is assumed to be sinusoidal in this chapter for

simplicity reason.
I.(z) = I_ sin k(h„ - z) (2.18)

2 2m 2

with k = 2-n/X (X = wavelength), and the parallel field component E „ due to an assumed

sinusoidal current distribution, I sin k(h - z) , on antenna #1 is given by

exp(-jk r^ ) exp(-jk r^)

^z21 = -J 30 1,^ { ^T - ^

exp(-jk r ) - 2 cos kh — ] , (2.19)

where h is the length of antenna #1, d is the horizontal separation distance between

antennas

,

and

r^2 2.1/2

r^ = [d + z ]

r^2 ,. ^2-,l/2

r^ = [d + (h^ - z) ]

r^2 ,^ N2n1/2 r^ = [d + (h^ + z) ]

Inserting (2.17) and (2.19) into (2.16), we obtain an expression for the mutual

lance. For the case of half-wave dip< a much simplified expression results [4]:impedance. For the case of half-wave dipole antennas (quarter-wave monopoles), h = h = \/H,

where

^21 = ^21 *

J ^21 ' ^^-^^^

2.2 2J/2

^2^ = 15{2Ci(kd) - Ci[(k'^d^ + n'^)^''^ - tt]

- Ci[(k^d^ + -n^)^^^ + tt]}

and

X^^ = 15{Si[(k^d^ + TT^)^^^ - U] + Si[(k^d^ + TT^)^^^ + ^]

-2Si(kd)}

Similar expressions may be derived for other antenna lengths [4]. The above sine

integral Si(x) and cosine integral Ci(x) are defined respectively as

X Si(x) = / ^i^^ dv , (2.21)

^ and

Ci(x) = - / ^^^-^ dv . (2.22)

V

X

The cosine integral Ci(x) may also be rewritten as:

Ci(x) = ]ln X + C - S^ (x) , (2.23)

where C = 0.577 is known as the Euler's constant, and
S (X) = / 1^-5°^ dv

^

2 4 6 ^ ^ • '^ ••• (2.24)

2*2! H'Hl 6'6!

Figure 2.3 shows the amplitude and phase of the mutual impedance between equal-length dipole

antennas as a function of antenna separation distance. More general cases of the mutual

impedance calculations for collinear or echelon arrangements with arbitrary antenna lengths

can be found in [5].

2.3 Measurement Technique and Results

The test range geometry and mutual- impedance relations have been shown in Fig. 2.1 for

horizontal polarization. The value of the base current, 1.(0), is calculated by assuming that one watt of power is delivered to the transmitting dipole (#1 antenna, P. = 1 W) which has an input impedance given in (2.2). In our calculation of Z , we have simplified the

procedure by neglecting Z and Z , thus assuming that the input impedance of the

transmitting dipole is unaffected by the current induced in the receiving dipole's finite load

impedance. For calculating Z in (2.6), we have neglected Z . The value of Z is

calculated using equations given by Schelkunoff [6]. The input impedance of the receiving

dipole, Z , is calculated in the same manner, and is then used to calculate the received

power according to (2.14). The site attenuation is then calculated using (2.15) with Z = Z o n = 100 n. We also have assumed that the site has a perfectly conducting ground (a = -1 for

horizontal polarization, and +1 for vertical polarization).

Since the relative insertion loss represents the ratio of the power delivered to the

transmitting dipole terminals to the power received at the receiving dipole terminals, it is a positive quantity. Site attenuation is the mini mum insertion loss occurring when the

receiving dipole is scanned in height. The height-scan patterns presented in Fig. 2.4 show

calculated insertion loss for horizontally polarized (HP) and vertically polarized (VP)

dipoles with a 3 m separation distance. The transmitting dipole is 2 m above perfectly

conducting ground. Site attenuation is the single minimum value of each pattern. Figure 2.5

shows site attenuation versus frequency for separation distance (d) of 3 m, 10 m, and 30 m. Heights scanned are 1 m to 4 m at the 3 tn and 10 m separation distances, and 2 m to 6 m at the

30 m separation distance except for vertical polarization when the half-length of the

receiving dipole would touch the ground [1]. The lower tip of the vertically polarized dipole

is always positioned 5 cm or more above the ground. Calculated site attenuation provides an ideal reference. Deviations of measured data from the reference indicate site imperfection.

Figure 2.6 shows the measurement procedure used for determining site attenuation. The

reference relative insertion loss obtained from Fig. 2.6(a) is a received signal level

expressed in dBm, dependent upon the output level of the signal generator and balun, cable,

and attenuator losses. The output level remains fixed during the measurement procedure and is

somewhat arbitrary. It is less than the maximum level acceptable by the receiver but great

enough so that the signal level in Fig. 2.6(b) is well above the ambient noise level.

The dipole antennas used to measure site attenuation at NBS have hybrid junctions for

antenna baluns. Equal length coaxial cables from the dipole terminals to the hybrid junction

form a lOO-fi, balanced shielded transmission line. As a result, the receiving and

transmitting system impedances are 100 fi. This value is used to calculate the reference site

attenuation data shown in Fig. 2.5, as the solid lines for very thin dipoles and (++) points

for dipoles with dimensions of those actually used for the measurements. To insure that this

impedance is approximately 100 fi, miniature 3 dB attenuators are permanently installed in the

four separate cables at the point of attachment to the dipole terminals.

The receiving dipole is moved up and down over the specified height range by a person

below the ground screen who observes the receiver display. The maximum received signal level

is recorded by a peak sample-and-hold circuit in the receiver. The magnitude of the

difference between the reference insertion loss value and this maximum measured value gives

site attenuation directly. No additional measurements or corrections are required. The

measured data are shown as the circles in Fig. 2.5.
2.M Error Estimate

Test range antenna heights and separation distance are set within ± 1 cm assuming the

ground screen surface is perfectly flat. Dipole height is measured at the dipole feedpoint.

The extensible dipoles droop about 16 cm at the tips at 30 MHz, No effort is made to keep

these dipoles straight. Calculated errors in site attenuation are at most ±0.09 dB as a result of possible positioning errors (neglecting dipole droop).

The stability of the receiver, signal generator, hybrid junctions, and cables combination

is determined by the repeatability of the reference insertion loss measurements. Fig. 2.6(a),

performed before and after each subset of measurements. (The three subsets of measurement

frequencies are determined by the frequency ranges of the three sets of hybrid junctions.)

Over these typically two-hour time periods, the difference between the initial and final

reference insertion loss data is at most ±0.11 dB. This variability appears to be

predominantly caused by cable handling, connector mating, and moving the signal generator.

Since this is a relative insertion loss measurement, not an absolute one, the primary

uncertainty is the manufacturer's specified "cumulative fidelity" for the receiver (spectrum

analyzer) of "i ±1.0 dB over to 80 dB display, 20-30°C". Therefore, a simple worst-case

error estimate is (±0.09 ±0.11 ±1.0) i ±1.2 dB.

The statistics of the measured data imply that the uncertainty may be less than half of

this worst-case estimate if it is assumed that the calculated data are correct. The average

difference between the measured and calculated [calculated - measured, dB] data for

horizontal polarization is -0.2 dB with a standard deviation of 0.4 dB calculated using the dB

values.

The data for the single measurement set upon which this chapter is based are shown in

Table 1 . This was the first set of data measured after the preliminary set used to determine

the suitability of the various test components. Failure of the air supported fabric cover

over the NBS ground screen facility has temporarily halted further measurements. The greatest

difference between measured and calculated site attenuation is 1.02 dB for horizontal

polarization, where the effect of feed cable reflections is negligible, and 1.94 dB for

vertical polarization.

REFERENCES

[1] FitzGerrell, R. G. Site attenuation. Symposium record of the 1985 IEEE Int. Symp. on

EMC; 1985 August; Wakefield, MA. 612-617. Also, IEEE Trans, on EMC; EMC-28(1): 1986

February.

[2] I.E.C. 12A (Secretariate) W, 1978 April.

[3] FCC Bulletin, OCE 44 (Docket No. 21371, Appendix B) , Calibration of a radiation

measurement site: site attenuation. U.S. Government Printing Office, Washington, D. C;

1977 September.

[4] Jordan, E. C. Electromagnetic Waves and Radiating Systems. Englewood Cliffs, N. J.:

Prentice-Hall; 1950.

[5] King, H, E. Mutual impedance of unequal length antennas in echelon. IEEE Trans.

Antennas Propag. , Vol. AP-5(3): 306-313; 1957 July.

[6] Schelkunoff, S. A.; Friis, H. T. Antennas, Theory and Practice. New York, N. Y.: Wiley

and Sons; 1957.

10
Anienna «1

TransmiUing A/2

dipole

11 / . /

' Anienna 1 I

/

Antenna #2

Receiving A/2

dipole

/

Antenna 2'

V

Figure 2.1. Two half-wave dipoles above a ground plane.

Image

Figure 2.2. Computation of mutual impedance.

11
o

I«»

*Co

Antenna spacing -rFigure 2.3. (a). Magnitude of the mutual impedance between

two equal-length dipole antennas.

V

o

co

•)o

Q.

Antenna spocing -rFigure 2.3(b) Argument of the mutual impedance between

two equal-length dipole antennas.

12
DIPOLE HEIGHT VS INSEi^TICN lOS3 ...mP FREQUENCIES, L TO R: 30, 97, 311, 1000 MHZ

lO.O IS.O 20.0 25.0 30.0 35.0 10,0 45.0 RELATIVE INSERTION LOSS, DB

50.0

01 POLE HEIGHT VS INSERTION LOSS ,VP

o

-REQUENCIES, L TO R: 44, 143, 459, 1000 MHZ

Z a

X

2o /

/

// ^J JT

o (

<

>

o

\ \

N

<z>

1 lO.O 15.0 20.0 2S.0 30.0 35.0 10.0 «S.O RELATIVE INSERTION LOSS, DB

50.0

Figure 2.4. Selected relative insertion loss data calculated

for horizontally polarized (HP) and vertically

polarized (V?) dipoles over perfect ground.

Separation distance is 3 tn; transmitting dipole

height is 2 m.

13
SITE RTTENURTION V5 FREQUENCY HP

o

QQ

o

m

SOURCE HEIGHT, 2.0 M

DI5T = 3, 10, & 30 M

VP

FREOUENCY, MHZ

Figure 2.5. Site attenuation for horizontally polarized (HP)

and vertically polarized (VP) dipoles over perfect

ground. Solid curves are calculated for half-wave

dipoles with radius of l.E-30 m; + + points are calculated for dimensions of actual dipoles.

Measured data are shown as o o

.

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16
Chapter 3. TRANSVERSE ELECTROMAGNETIC (TEM) CELLS

One of the main limitations inherent in most EMC/EMI susceptibility tests is the

requirement of an antenna to radiate electromagnetic waves. Typical antennas are bandwidth

limited, and do not have linear phase response versus frequency. Hence, they are useful

primarily for frequency-domain measurements, and their applications to transient, impulsive

EMI testing and evaluation are very limited. In addition, for accurate measurements, the

separation distance between the source antenna and equipment under test (EUT) should be

sufficiently large to ensure far-field conditions. This is not always possible, especially in

confined chambers or enclosures. Quite often, the requirements of sensitivity and a highlevel testing field dictate the need for near-field measurements, which introduce other

problems related to field uniformity over the test volume occupied by the EUT and interaction

effects. Some of these difficulties may be eliminated or minimized by using TEM cells,

because they themselves serve as the transducer and thus eliminate the use of antennas.

The TEM cell was designed based on the concept of an expanded transmission line operated

in a TEM mode. As shown in Fig. 3.1, it consists of a section of rectangular coaxial

transmission line (RCTL) with tapered sections at both ends. The taper is used as a transition to match the RCTL to standard 50-Q, coaxial cable connectors at the two ports of the

cell. The taper sections should be gradual and long enough to minimize perturbation of the

TEM wave as it passes from one section to the other. It is generally recommended that this

length be at least one half of the cell's width. One of the two ports is usually terminated

with a 50-f2 load, while the other port is connected to either a source or a receiver,

depending on whether the cell is used for radiated susceptibility or emission testing. The

TEM cell provides a shielded environment without introducing the multiple reflections

experienced with a conventional shielded enclosure [1]. Hence, external EM signals will not

affect the measurement of low-level radiated emission from the EUT. Alternately, a high-level

test field generated inside a cell for performing radiated susceptibility tests will not

interfere with external electronic systems.

To support a TEM mode, the cell is necessarily a two-conductor system with the region

between the inner and outer conductors (either upper or lower chamber) used as the test zone.

The cross section of a TEM cell is depicted in Fig. 3.2. Although the center septum (inner

conductor) is normally designed to be midway between the top and bottom outer conductors, its

position can be modified to have a vertical offset to allow a larger test zone provided by one of the chambers [2], [3]. We limit our discussions in this chapter only to cells with

symmetrically located inner conductors, however.

TEM cells offer several advantages in performing EMC/EMI measurements of electrically

small equipment and devices. They are portable, simple to build [M], useful for broadband

swept-f requency measurements, and capable of providing test field strengths from a few

microvolts per meter to a few hundred volts per meter. The cost to build a TEM cell is much

lower than those facilities such as anechoic chambers and shielded enclosures. The TEM field

generated between the inner and outer conductors simulates very closely a planar far field in

free space, and has constant amplitude and linear phase characteristics [5].

The application of TEM cells has limitations. The usable frequency range is from dc to

an upper limit determined by the appearance of high-order modes [2], [6], [7]. The volume

available for testing purposes is inversely proportional to the upper frequency limit. In addition, since the EUT is placed at the center of the test zone, its size should be small

relative to the test volume in order that the field structure associated with the ideal TEM

mode existing in an empty cell will not be significantly perturbed.

The most important aspect for a TEM cell user to recognize is that the results obtained

from measurements made inside a cell have to be interpreted correctly. The characteristics of an EUT inside a TEM cell are different from those in other environments such as free space

[8]. Thus interpretation is crucial, especially if a meaningful comparison with other

measurement methods is attempted.

Before presenting the details of using a TEM cell to make radiated susceptibility and

emission tests, we will first summarize some basic TEM cell properties.

17
3.1 Impedance

The characteristic impedance, Z , of an RCTL may be expressed in terms of the distributed

capacitance per unit length of the transmission line, C , by [2], [8]:

Z = n e /C (3.1) o o o o where n

= 120ir is the free-space intrinsic impedance in ohms, and e = (10) /36it is the air

permittivity in farads per meter.

Determination of C may be achieved by the method of conformal transformation whereby the

geometry of Fig. 3.2 is transformed into a simpler configuration whose capacitance is already

known. Because of symmetry, the total capacitance per unit length is just twice that due to

either the upper or lower chamber. It can be expressed exactly in terms of the Jacobian

elliptical function, which has to be computed numerically [8]. If, however, the aspect ratio

satisfies the condition b/a i 1, an approximate but simpler expression may be obtained [8]:

C /e = 4(w/b) + - in{] + coth ^ ) - AC/e , (3.2)

IT 2b

where the cell parameters a, b, g, and w, all in meters, are indicated in Fig. 3.2. In this

form, it is easy to identify the first term, Mw/b, as the plate capacitance between the center

septum and the horizontal walls (top and bottom outer conductors), the second term involving

the gap parameter g as the fringing capacitance between the edge of the septum and the

vertical sidewalls, and the third term AC/e as a correction needed to account for the

o interaction between the two edges. Under the practical condition, w/b 2 1/2, AC is

negligible. A graphical presentation for C /e is given in Fig. 3.3.

If a characteristic impedance value of, say, 50f2 is desired, we require C /e = 7.5M, A number of possible TEM cells may be designed, based on (3.2) or Fig. 3.3, to meet this

requirement.

3.2. Highei—Order Modes

As mentioned before, the upper frequency limit for a TEM cell is determined by the

appearance of higher-order modes which perturb the desired TEM-mode field distribution

[2],[6],[7] ,C9]-[13]. For an infinitely long central RCTL section, the cutoff wavelengths for

the first few higher-order modes may be determined numerically [10]. Figure 3.^ presents

results for various cell configurations. Alternately, if the gap is small such that terms of

2 2 the order (TTg/2a) and (k g) can be neglected when compared to unity, the modal equations

listed in Table 3.1 may^ be used to find the cutoff wavenumbers k and frequencies f of the

c ^ c first few higher-order modes [13].

In reality, the cell is finite in length and the tapers cause it to act as a cavity. The

30i by

resonant frequencies f associated with a particular mode of cutoff frequency f are given

fpes = ^ ^c

" (PC/2cl)^ ]^^^

. (3.3)

Q Where p is an integer, c = 3(10) m/s is the speed of light, and d (in meters) is the resonance length. Because of the tapered sections, d is not well defined. An average

"overall cell length" is usually taken as a first approximation [6]. However, for best

results the resonant length needs to be determined mode by mode and cannot be specified

theoretically. Some empirical data do exist [12], [13].

It is important to note that a) the influence of the first few higher-order TE modes does

not become significant until approaching a resonant frequency, and b) if the septum of the

cell is centered symmetrically, the odd-order TE modes are not excited in the empty cell

(these modes may exist when an EUT is placed in the cell). Thus the upper useful frequency

18
can exceed the multimode cutoff frequency of the first higher-order mode, but should be less

than this mode's associated resonant frequency. For example, the cutoff and resonance frequencies for a cell with a = b = 1 m, w/a = 0.8 (or g/a = 0.2), and d = 4 m are approximately M3 MHz and 56 MHz. Such a cell should be useful at frequencies up to

approximately 50 MHz. However, because the TEM cell is a high-Q cavity, the resonances appear

at sharply defined frequencies. Thus, there may well exist windows between resonances where

TEM cell usage is still quite valid. This very much depends on the particular cell

application [12].

Efforts have been made to extend the use of TEM cells to frequencies above cutoff by

installing RF absorbers inside the cell [11]. While this effort helps to lower the Q factor

of the cell and suppress the multimoding effects, it also has some effect on the fundamental

TEM mode. Thus care must be exercised when considering the placement of absorbing materials

inside a TEM cell.

3.3 Field Distribution

An analytical expression for the electric field inside an empty TEM cell over the cross section shown in Fig. 3-2, when operated in a TEM mode, is also available in terms of Jacobian

elliptical functions [2], [8]. Numerical results for the electric field in a typical symmetric

cell are presented in Fig. 3.5. The results have been normalized with respect to V/b, which

represents the electric field at the center of the test zone (x = 0, y = b/2) where

V = (P Z

)^''^ (3.4)

n o

is the voltage in volts between the inner and outer conductors, P the net power in watts

flowing through the cell, and Z the characteristic impedance in ohms given in (3.1).

Normalized results in the form of x and y components are given in Tables 3-2 and 3.3- Field

distribution for certain higher-order modes, while not the major concern of TEM cell users,

may be found elsewhere [2], [1 2], [1 3]

.

3.4 Performing Radiated Susceptibility Tests

The main purpose of radiated susceptibility testing is to determine if and how EM energy

is coupled into the EUT so as to cause degradation to the equipment's performance. Thus, a criterion for what constitutes degradation of the EUT and how this is translated into

measurable parameters is normally established by the user. This criterion may be in the form

of video or audio indicators, or by other means when the EMI coupling exceeds a predetermined

threshold.

The following steps are suggested as a systematic approach for making the radiated

susceptibility evaluation [14].

Step 1. Place the EUT inside the TEM cell: The first step is to place the EUT in a TEM

cell, centered in the lower half space below the septum. The first position (position A) as shown in Fig. 3.6a is near the floor but insulated from the floor with approximately 2 cm of

foam dielectric. Plastic foams with dielectric constants of 1.04 and 1.08 are readily

available, are almost invisible electrically, and make good supporting material. If grounding

of the equipment case is desired, the EUT would then be placed on the floor. This position

(position A) is used to minimize exposure of the input/output leads associated with an EUT to

the test field (to be explained in step 2). Another common EUT position (position B) for

testing, as shown in Fig. 3.5b, is midway between the septum and the floor. Again, the EUT is

supported on a foam material of low dielectric constant. This position (position B) increases

the exposure of leads to the test field. A comparison of the test results to be taken later

for both positions A and B should give some indication of how energy is coupled into the EUT.

After placing the EUT in positions A and B, the EUT may be reoriented as desired, relative to

the field polarization. Typically, the first orientation is with the EUT lying flat as in

normal use. Care must be taken to record the placement location and how this is done so that

it can be repeated if necessary. It may be helpful to mark the bottom of the cell with a uniform array of scribe marks to assist in determining placement locations precisely.

Step 2. Access the EUT as required for operation and performance monitoring: The EUT

input/output and ac power cables should approximate those anticipated for use. Cables should

be the same length if possible, be terminated into their equivalent operational impedances so

19
as to simulate the EUT in its operational configuration, and be carefully routed inside the

cell to minimize field perturbation. Dielectric guides or holders may be installed in the

cell to assure repeatability of the placement location of the cables. These may be placed on

the floor to allow the cables to be covered with conductive tape (minimum exposure) and/or on dielectric standoffs to provide coupling of the test field to the leads. If required, any excess portion of the EUT's leads (wiring harness) may be carefully coiled and covered with

conductive tape on the floor of the cell. When the leads are bundled together, it may be

helpful to twist the input/output monitor leads as separate conductor pairs or use shielded

cables to minimize cross coupling between them. It may also be necessary to space the

windings in the coil to avoid introducing resonances associated with the coil inductance and

distributed capacitance. If braided RF shielding is used, it should be placed in electrical

contact with the cell floor, and not in contact with the case of the EUT unless a common ground between the EUT and cell is required. Grounding the two together will influence the

results of the susceptibility measurements. The input and output leads, after being connected

to the appropriate feedthroughs for accessing and operating the EUT, should also be filtered

to prevent RF leakage from the cell, otherwise the shielding integrity of the measurement

system will suffer. Care must be exercised in selecting these filters so they do not

significantly affect the measured results. The monitor leads used for sensing and telemetering

the performance of the EUT may require special high-resistance lines made of carbonimpregnated plastic or fiber optic lines to prevent perturbation of, or interaction with, the

test environment. DC signals or signals with frequency components below 1 kHz may be monitored

via the high-resistance lines. Radio frequency signals should be monitored via fiber optic

lines.

If the monitor signal is at a frequency or frequencies sufficiently different from the

susceptibility test frequency or frequencies, metallic leads may be used with appropriate

filtering (high-pass, low-pass, band-pass, etc.) at the bulkhead. Such leads, however, will

cause some perturbation of the test field; thus, their placement location must be carefully

defined for future reference. Note that a separate, shielded filter compartment should be

provided on the outside of the cell for housing the filters, as shown in Figs. 3-6a and b.

Step 3. Connect the measurement system as shown in Figures 3.7 and 3-8: Figures 3.7 and

3.8 show the block diagrams of systems using the TEM cell for susceptibility measurements.

These figures are used for frequencies from approximately 10 MHz to the recommended upper

frequency for the particular cell used. At frequencies below 10 MHz, the dual directional

coupler and power meters are replaced by a voltage monitor tee and RF voltmeters. Figure 3.7

is a diagram of essentially a discrete (manually operated) system or can be used for swept

frequency testing. Figure 3.8 is a diagram of a system for automated testing under computer

control, which allows the test field level in the cell to be carefully controlled and

progressively increased over selected frequency ranges and intervals while monitoring the EUT

performance. If degradation occurs as determined from a pre-established threshold limit and

as evidenced by the EUT monitors, the computer can respond interactively with the EUT, thus

limiting the test field level and preventing damage to the EUT. The computer can also be used

to store the raw data, process the data incorporating correction factors as needed and output

the results to printers or plotters according to the software instructions and format.

Step 4. Initialize the measurement system: This includes zeroing the appropriate

instrumentation and measuring the residual offset values of the EUT monitors with the RF source turned off and the EUT turned on in the desired operation mode. These values are then

recorded for future reference.

Step 5. Establish the test field and determine the EUT response: After initialization

of the measurement system, the RF source is then turned on at the desired test frequency,

modulation rate, test wave form, etc., and its output level is increased gradually until the

maximum required test level is reached or the EUT response monitors indicate vulnerability.

Care must be exercised to ensure that sufficient time is spent at each frequency and field

level to allow the EUT to respond. The EUT susceptibility profile is then determined for each

position (A or B as shown in Figs. 3.6a and b) and orientation. It may be necessary to test

all three orthogonal orientations of the EUT inside the cell. This is required if all

surfaces of the EUT to be tested are to be polarization matched to the TEM field of the cell.

If the test frequency is below 10 MHz, the electric field level in V/m generated inside

the cell is determined by the RF voltmeter reading, V „ in volts, in accordance with V ./b, ° rf rf

where b is the separation in meters between the septum and the floor. When the test frequency

is 10 MHz or above, where the electric length of the cell is significant, the electric field

20
level is determined by V/b, where V is given in (3.4) and the net power, in watts, may be

determined by

P^ = C-P. - C^P^ (3.5)

n II r r

with C„ and C as the respective forward and reverse coupling ratios of a calibrated bidirectional coupler, ^ and P. and P as the indicated incident and reflected coupler sidearm

1 r power readings in watts. Note that the absolute level of the test electric field inside the

cell is a function of the location of the EUT in the test zone. An appropriate correction can

be made based upon the particular cell's cross section. Note also that the size of the EUT

relative to the test volume can influence the determination of the amplitude of the test field

[see Sect. 3.5].

If the objective of the measurement program is simply to reduce the vulnerability of the

EUT to EMI without the additional requirement of determining worst-case susceptibility as a function of absolute exposure field level, one EUT orientation with input/output lead

configuration may be tested in one particular operational mode under a pre-selected

susceptibility test-field waveform and level. Similar tests may then be duplicated at the same test position with the same lead configuration and test-field waveform and level, after

the corrective measures such as providing additional shielding, etc. are made to the EUT.

These testing results are then compared to determine the degree of improvement.

Sometimes, it is desirable to monitor the field distribution inside the cell using small

calibrated electric and/or magnetic probes, while an EUT is in position. If this is the case, one must be careful in interpreting these monitored results, because the results are a combination of the incident TEM field launched inside the cell and the scattered fields from

the EUT and its leads in the near-field. The field so monitored can be quite different from

the unperturbed test field, leading to potentially erroneous conclusions. Whenever possible,

it is preferable to mount the field-monitoring probes in the other half space of the cell in

the mirror image location of the EUT in order to avoid the interaction between the probe and

EUT.

3.5 EUT Scattering in a TEM Cell

As indicated earlier, a test object will scatter the incident TEM mode. If the EUT is

kept small compared to the chamber height and the operating wavelength, it is reasonable to assume that the cell environment will not be significantly perturbed from its empty state.

Small obstacle theory, the dual of small aperture theory, may be used to investigate this

problem in a more formal manner [15]. When only the TEM mode is considered, the cell may be

viewed as a simple transmission line circuit. The EUT introduces a load which may be

represented and analyzed by an equivalent T-network. This allows us to investigate the

effects of the load perturbation on the overall TEM cell transmission line characteristics,

such as the input impedance. The excitation of the initial higher-order modes may also be

studied. This enables one to theoretically assess the expected field perturbation due to EUT

scattering. These two topics are discussed briefly below. A more detailed discussion may be

found in [15].

The T-network representing the EUT loading may be included in the TEM cell transmission

line circuit as shown in Fig. 3.9. The EUT is assumed to be centered in the central test

section of length 2Lpp„ . Ideally, the tapered sections of length L„.pp,o will have the same

characteristic impedance (50 Q) as the test zone, i.e. Z^„„. = Z„._,„„ = Z^. In practice there

KL i L 1 Ar CjK U is some deviation, thus the taper sections are shown explicitly in the circuit model. Let a- and b-^ represent the forward and backward scattering coefficients for the TEM mode. Explicit

expressions depending on the polarizabilities of the EUT may be found in [15]. Small obstacle

theory predicts that the load impedances Z and Z (see fig. 3.9) will be given by

3. D

^a ^oh0^2 +. a^ - b^'

(3.6)

21
If the TEM mode is weakly scattered (a^ and b. small), then Z and Z are approximately given

U U 3, D by

z = -z

,^0 -

'^o, a 0^ 2

Z, = -Z,

(3.7)

b O'a^ . b^^

As the EUT size vanishes, both a„ and b^ tend toward zero. Therefore Z ->• (short circuit)

a and Z - oo (open circuit) as expected.

Tests were performed in a typical NBS cell to assess the validity of the above circuit

representation. The cell has dimensions a = 15 cm, b/a = 1, and g/a = .17, yielding a theoretical characteristic impedance Z„ of 51.6 fi. Cell impedances were also measured with a time-domain r ef 1 ectometer . This technique yielded Z„._,„„ ~ 51 Q and Z„^„, = M9.5 Q.

Agreement between the theoretical impedance computation, actual measured impedances, and the

desired 50-fi value is usually considered acceptable if they are within 2 Q. In the figures

that follow, the measured impedance values are used in the circuit model. The central section

is 30 cm long (L„„„, ) while each of the tapers is 25 cm (L„.„„^) in length. Figure 3.10 shows

the measured and computed (based on fig. 3.9) input impedances (magnitude) for the empty cell

terminated with a 50-n load. At low frequencies (< 100 MHz), the well-defined 50-fi load

impedance dominates. As frequency is increased, the taper and RCTL impedances become

important and the mismatches begin to cause standing-wave variations from the load value.

Theory agrees well with the measured data below 400 MHz.

Figure 3.11 shows input impedance data for the same TEM cell loaded with a conducting

sphere. The sphere diameter is ^3% of the test chamber height. The theoretical curve is

little perturbed from the measured data even though this test object occupies a significant

portion of the chamber. In fact, both of these curves are similar to the empty-cell data.

This type of data suggests that the impedance characteristics of the cell are not highly

sensitive to the presence of an EUT.

The scattering coefficients for higher-order modes in a TEM cell may be worked out in a manner similar to those for the TEM mode (a_,b_). This enables us to study the field

perturbation due to the EUT presence. Rather than consider any specific position in the cell

we will look at the primary TEM mode field components (E ,H ) at the test chamber center

(x=0,y = b/2) as a function of z, the longitudinal distance from the EUT. Figure 3.12 shows

data for the same cell and sphere as tested in the previous figures. The frequency was chosen

such that k_a = 1.4 which is very near the cutoff frequency of the first higher-order mode.

The scattered TE.. and TE modes are included in addition to the scattered TEM mode. The

TE^ mode does not significantly contribute for a centrally located EUT. As may be seen,

there is greater perturbation in the backward direction (z < 0) versus the forward (z > 0).

This results from: 1) |b_| > la.], and 2) the phase beating between the reflected and incident

TEM mode. If the contributions from the individual modes are considered separately, it turns

out that the scattered TEM mode accounts for almost all the perturbation. In the backward

direction the electric field is reduced while the magnetic field is increased resulting in a decrease in the wave impedance in the cell. One implication of this data is that if a field

probe is used to estimate the strength of the incident TEM mode, it should be located forward

of the EUT for best results. This figure represents somewhat of an extreme case since the EUT

is large and the operating frequency is near the upper limit of the cell. Even so, the

deviation in the primary field components is less than ^0%.

These examples indicate that the TEM cell environment is not greatly affected by

reasonably sized test objects. Smaller spheres would have yielded correspondingly smaller

22
variations. For EUTs of different shapes scattering could possibly be accentuated somewhat

due to corners, points, etc., but the same conclusions should hold true. Experience and

theory suggest that limiting major EUT dimensions to less than one third the test chamber

height should avoid significant perturbations.

3.6 Performing Radiated Emission Tests

Electronic or electromechanical equipment may emit energy which interferes or interacts

with the normal operation of either the system and/ or other receptors. To ensure the EMC of

such systems, it is important to determine quantitatively their radiation characteristics.

Since a TEM cell is a reciprocal device, it can be used to detect the emission too.

Equipment not designed as a radiator represents unintentional leakage sources which may

be considered electrically small. As such, the leakage currents on the exterior surfaces of

the equipment, treated here as an EUT, can be modeled as equivalent electric and magnetic

short dipole sources [16]-[18]. These dipole sources may then be combined vectorially to form

a composite equivalent source consisting of three orthogonal electric and three orthogonal

magnetic dipole moments as represented in Fig. 3.13, each having an amplitude and a phase.

When an unknown source object (EUT) is placed at the center of a TEM cell, its emission

couples into the fundamental TEM mode and propagates toward the two ports of the cell. With a hybrid junction inserted into a loop connecting the cell outputs as shown in Fig. 3.1^, we are

able to measure the sum and difference powers and the relative phase between the sum and

difference outputs. This way of measuring the relative phase is very advantageous because it

avoids the complication of having to establish an absolute phase reference physically

connected to the EUT. Systematic measurements of the powers and relative phases at six

different EUT orientations are sufficient to determine the amplitudes and phases of the

unknown equivalent component dipole moments as depicted in Fig. 3.13, from which the

corresponding detailed radiation pattern and total power radiated by the unknown source in

free space can then be computed [16],

For the purpose of describing the source (EUT) position and related experimental

procedures, it is necessary to establish a coordinate system (x,y,z) for the TEM cell with the

origin at the geometric center of the inner conductor. The unknown source EUT may be placed

at (0, y , 0) . We assign another coordinate system (x', y', z') with respect to the center of

the EUT, as shown in Fig. 3.15.

Initially, we align x-x',y-y', and z-z' as shown in Fig. 3.15a. The first measurement

orientation is obtained by rotating the EUT counterclockwise by an angle of Tr/4 about the z'-

axis so that its position relative to the TEM cell is shown in Fig. 3.15b. We measure the sum and difference powers in watts and the relative phase in degrees between the sura and

difference outputs, and designate them respectively as P , P^, , and cj) . We then rotate the

EUT by an additional Tr/2, also counterclockwise about the z'-axis as displayed in Fig. 3.15c

to the second measurement orientation. We measure the sum power, difference power, and the

relative phase between them as P „, P „, and (()„.

We next align the coordinate frames such that x = y', y = z', and z = x', as shown in

figure 3.16. Then we rotate the EUT counterclockwise by an angle of ir/^ about the x'-axis

serving as the third measurement orientation, take the sum power P , difference power P^n.

and the relative phase between the sum and difference outputs c})^. Rotating the EUT

counterclockwise by another ir/2 about the x'-axis and taking the same sequence of measurements

yield P^^, P^^, and <|,^.

Finally, we align the coordinate frames in accordance with x = z', y = x', and z = y', as shown in figure 3.17, rotate the EUT in a similar manner about the y'-axis, and measure P ^,

si? Pd5' S' ^s6' Pd6' ^"^ *6-

After collecting the measured sum and difference powers and applying the Lorentz

reciprocity theorem, we obtain the following [18]:

23
m^ = (P , + P , - P^. - P,. + P,. + P^.)/(2q^) (3.8) ex si s2 s3 sM s5 so

m^ = (P 1 + P.n + P,^ + P.^ - P^p; - P,ft)/(2q^)

' (3.9)

ey si s2 s3 s4 s5 sb

m^ = (-P , - P ^ + P ^ + P , + P ^ + P .)/(2q^) (3.10) ez si s2 s3 s4 s5 so

m^ = (Phi ^ Phq - P^^ - Ph^ * Phc; ^ PH^)/(2k^q^) (3.11) mx dl d2 d3 dM d5 do

m^ = ^Phi * P^o ^ Ph^ * Phu - Ph^ - P.A^/^^k^q^) (3-12)

my dl d2 d3 d'4 d5 do

m^ = (-P.. - P., + P,^ + P., + P.. + P..)/(2k^q^) ; (3-13) mz dl d2 d3 d4 db do

m m COS0 , = (P , - P _)/(2q^) (3.14) ex ey el si s2 ^

m m COS0 _ = (P _ - P „)/(2q^) (3.15)

ey ez e2 s3 s4

m m COS0 _ = (P ^ - P .)/(2q^) (3.16) ez ex e3 s5 s6 ^

m m cose . = (P,_ - P..)/(2k^q^) . (3.17) mx my ml d2 dl

m m COS0 . = (P^,, - P^-)/(2k^q^) (3.18)

my mz m2 d4 d3

m^ m ^ cos0^_ = (P.. - P_)/(2k^q^) ; (3-19) mz mx m3 do d5

B ^ = ^ - ^l) , (3.20)

e3 ez ^ex

, = \|j - 4j ; (3.21 ) m3 mz mx

and q is the normalized amplitude of the vertical electric field which would exist at the

center of an empty TEM cell when it is operated in a receiving mode and is excited by an input

power of 1 W at one end and terminated at the other end with a matched load. Thus, q = 1/2 1/2

(50) /b in Q. /m, which is determined by the height of the cell.

In obtaining (3.8) through (3.19), we have assumed that the frequency of the unknown

interference source (EUT) to be detected by the spectrum analyzer included in the measurement

instrumentation as shown in Fig. 3.1^ is such that it allows only propagation of the dominant

mode inside the cell. The size of the interference source to be tested must be small relative

to the test volume of the cell. This is required to minimize the potential perturbation to

the field distribution inside the cell, as was cautioned before. Under the above assumptions, we see from (3.8)-(3.10) that the amplitudes, m ., i = x, y, z, of the unknown component

electric dipole moments in ampere-meters are obtainable by the sum powers only, and from

(3.11)-(3.13) that the amplitudes, m . , of the unknown component magnetic dipole moments in

ampere-square meters are obtainable by the difference powers only.

Once m . and m . are determined , they can be used to compute the total power, in watts,

ei mi 'J F f > radiated in free space by the same source as

here

^el = ^ex ^ > ey Qe2

= '^ey ^ez

%1 = ^mx ^my Qm2 = ^my ^mz

24
2

P = ii2^ [m^ + m^ + m^ + k^(m^ + m^ + m^ ) ] (3.22)

T 2 ex ey ez mx my mz w / A

Note that if one is interested only in determining the total power radiated by the

2 2 source, which may^ be the case in practice, ^ ' all we need to extract are m . and m . based on the

ei mi

measured powers.

The corresponding far-field radiation pattern in free space depends, however, also on the

relative phases among the component dipole moments of the same and mixed kinds. The former

(same kind) can be extracted by (3. 1 ^)-(3. 1 9) with the constraining conditions that 10 . = ei

and Z0 . = 0, while the latter (mixed kind), \i) . - ili ., i,j = x,y,z but i ^ j , is much more

mi ^ei ^mj •^ complicated, involving the component dipole moments determined in (3.8)-(3.13) and the

measured relative phases, (j). , i = 1 , 2, ...,6, between the sum and difference outputs. This

is the reason why these relative phases are also measured and recorded. Detailed procedures

of extracting ^ . - \b . may be found elsewhere [18]. '^ ^ei ^mj •'

Note, however, that if one suspects beforehand that the unknown interference source may

be characterized by one kind of dipole moment only (either electric or magnetic), the relative

phase measurement will not be required [18]. In addition, if the source is believed to be made

of electric dipole moments only, then the sum power measurements will suffice to determine the

radiation characteristics of the source. Similarly, the difference power measurements will be

the dominant ones if the source is essentially magnetic dipoles.

Once the total power radiated in free space by the unknown source is determined as in

(3.22), the equivalent electric field strength, in V/m, due to the same source in free space

at a distance d will then be:

E = 5.477(P^)^^^/d (3.23)

Naturally, when the measured powers and phases are not contaminated by background noise or other inaccuracies, the source parameters so extracted and the radiation characteristics so computed are accurate and proven to be unique [18]. In the practical world, however, the

experimental data are always degraded somewhat by background noise, equipment limitations, and

the reading accuracy. These measurement imperfections will cause uncertainties in the deduced

results. A report giving necessary mathematical derivations for, and performing the analysis

of, these uncertainties is available [19].

The method outlined above has been successfully tested by a simulated example and two

practical examples, one with a known small spherical dipole representing the electric type

[16], [18], and the other with a known small loop antenna representing the magnetic type [20].

The deduced radiation characteristics such as the strength of dipole moment and the total

power radiated by the spherical dipole in free space agree very well with the measured results

obtained by other means [21].

Quite often, the equipment performance degradation or failure is also dependent upon the

interfering signal waveform. For this case, time-domain analysis may be required. TEM cells

can be used to satisfy this need too. For example, the block diagram shown in Fig. 3.18 may

be applied with the oscilloscope either connected directly to the cell measurement port, or with the oscilloscope connected to the predetection or postdetection outputs of the receiving

instrument. The latter arrangement using a receiver with the oscilloscope provides greater

measurement sensitivity. In either arrangement, the oscilloscope must be synchronized with

either the periodic detected signal from the cell or with an EUT-moni tored periodic signal

represented by the dashed line from the EUT. The measurement results can then be recorded by

photographing the oscilloscope display. If the emission is random in nature, the oscilloscope

cannot be synchronized properly and the detected signal must be either recorded with a video

disk or tape recorder to be played back frame by frame for analysis, or analyzed statistically

using amplitude probability distribution analyzers, etc.

3.7 Examples of Some TEM Cell Applications

TEM cells have been used for a number of applications. Initial work involved the

development of a cell to establish high-level fields for biological effects research [22].

25
During the early development, it was realized that its broadband frequency response made it a prime candidate for use in TEMPEST testing of EUT [23] and as an EM pulse simulator [2M]. The

largest known TEM cell in existence (8m x 20m x 2%) is located at Sandia Laboratories, in

Albuquerque, New Mexico, and it is used as a dual purpose facility for both EM pulse testing

and CW susceptibility testing of whole weapon systems. NBS recognized the cell's potential

for use as a calibration tool for establishing standard TEM fields. Considerable development

and theoretical work were done at NBS to carefully evaluate various TEM cells for this purpose

[6], [25]. TEM cells have also been used extensively for radiated susceptibility testing of

components by the automotive industry, as evidenced by the formal adoption of TEM cell testing

methods by the Society of Automotive Engineers (SAE) in the frequency range of ^^ kHz to 200

MHz [26]. One motor vehicle manufacturer has constructed a very large cell with a test volume

of 2m X 5m x 7m for whole-vehicle testing [27]. Another large cell, 2.8m x 2.8m x 5.6m, is in use at AT&T Information Systems for measuring both susceptibility and emissions of

communication equipment [28], [29]. More recently, 2m x 2m x Mm TEM cells have been evaluated

and proposed for use by the Electronics Industries Association (EIA) to measure TV/VCR

immunity to EMI [30], [31]. Finally, a recent application is the use of a pair of TEM cells,

called the dual TEM cell (one cell on top of the other), with a common aperture created

between them to evaluate the shielding effectiveness of materials [32], [33].

REFERENCES

[1] Free, W. R. Radiated EMI measurements in shielded enclosures. Proc. IEEE Int. Symp. on

EMC; 1967 July; Washington, D. C. M3-53.

[2] Tippet, J. C. Modal characteristics of rectangular coaxial transmission line. PhD

dissertation, Dept. of Electrical Engineering, Univ. of Colorado, Boulder, CO. 1978.

[3] Crawford, M. L. ; Workman, J. L. Asymmetric versus symmetric TEM cell for EMC

measurements. Proc. IEEE Symp. on EMC; 1978 June; Atlanta, GA. 20^-208.

[M] Decker, W. F. ; Wilson, W. A.; Crawford, M. L. Construction of a large transverse

electromagnetic cell. Nat. Bur. Stand. (U.S.) Tech. Note 1011; 1979 February.

[5] Crawford, M. L. Generation of standard EM fields using TEM transmission cells. IEEE

Trans, on Electromag. Compat., EMC-1 6(no.4) : 189-195; 197^ November.

[6] Tippet, J. C.; Chang, D. C.; Crawford, M. L. An analytical and experimental

determination of the cutoff frequencies of higher-order TE modes in a TEM cell. Nat.

Bur. Stand. (U.S.) Int. Rep. 76-841; 1976 June.

[7] Tippet, J. C.; Chang, D. C. Higher order modes in rectangular coaxial line with

infinitely thin inner conductor. Nat. Bur. Stand. (U.S.) Int. Rep. 78-873; 1978 March.

[8] Tippet, J. C; Chang, D. C. Radiation characteristics of dipole sources located inside

a rectangular, coaxial transmission line. Nat. Bur. Stand. (U.S.) Int. Rep. 75-829;

1 976 January. ...

[9] Mittra, R.; Itoh, T. A new technique for the analysis of the dispersion

characteristics of microstrip lines. IEEE Trans, on Microwave Theory & Tech., MTT19(1): 47-56; 1971 January.

[10] Gruner , L. High order modes in rectangular coaxial waveguides. IEEE Trans on Microwave

Theory & Tech., MTT-15(8): 483-485; August 1967.

[11] Crawford, M. L. ; Workman, J. L. ; Thomas, C. L. Expanding the bandwidth of TEM cells

for EMC measurements. IEEE Trans, on Electromag. Compat., EMC-20(3): 368-375; 1978

August.

[12] Hill, D. A. Bandwidth limitations of TEM cells due to resonances. J. Microwave Power,

18(2): 181-195; 1983.

[13] Wilson, P. F. ; Ma, M. T. Simple, approximate expressions for higher-order modes and resonances in TEM cells. IEEE Trans, on Electromag. Compat., EMC-28(3): xxx; 1986

August.

26
[31

Crawford, M. L. Improving the repeatability of EM susceptibility measurements of

electronic equipment when using TEM cells. SAE Technical Paper Series 830607, SAE Int.

Congress and Exp.; Detroit, MI. 1983 Feb. 28 - Mar. 4.

Wilson, P. F. ; Ma, M. T. Small aperture analysis of the dual TEM cell and an investigation of test object scattering in a single TEM cell. Nat. Bur. Stand. (U. S.

)

Tech. Note 1076; 198^1 October.

Sreeni vasiah, I.; Chang, D. C; Ma, M. T. Characterization of electrically small

radiating sources by tests inside a transmission line cell. Nat. Bur. Stand. (U.S.)

Tech. Note 1017; 1980 February.

Sreenivasiah, I.; Chang, D. C; Ma, M. T. Emission characteristics of electrically

small radiating sources from tests inside a TEM cell. IEEE Trans, on Electromag.

Compat., EMC-23(3): 113-121; 1981 August.

Ma, M. T. ; Koepke, G. H. A method to quantify the radiation characteristics of an unknown interference source. Nat. Bur. Stand. (U.S.) Tech. Note 1059; 1982 October.

Ma, M. T. ; Koepke, G. H. Uncertainties in extracting radiation parameters for an unknown interefence source based on power and phase measurements. Nat. Bur. Stand.

(U.S.) Tech. Note 1 06i< ; 1983 June.

Koepke, G. H. ; Ma, M. T. A new method for determining the emission characteristics of an unknown interference source. Proc. 5th Symp. Tech. Exhibition on EMC; 1983 March;

Zurich, Switzerland.

Crawford, M. L. ; Workman, J. L. Predicting free-space radiated emissions from

electronic equipment using TEM cell and open-field site measurements. Proc. IEEE Int.

Symp. on EMC; 1980 October; Baltimore, MD. 80-85.

Skaggs, G. A. High frequency exposure chamber for radiobiological research. NLR Memo,

Rep. 2218; 1971 February.

Crawford, M. L. Measurement of electromagnetic radiation from electronic equipment

using TEM transmission cells. Nat. Bur. Stand. (U.S.) Int. Rep. 73-306; 1973 February.

Pollard, N. A broadband electromagnetic environments simlulator (EMES). IEEE Int.

Symp. on EMC; 1977 August; Seattle, WA.

Crawford, M. L. Generation of standard EM fields for calibration of power density

meters, 20 kHz to 1000 MHz. Nat. Bur. Stand. (U.S.) Int. Rep. 75-804; 1975 January.

Society of Automotive Engineers (SAE) Recommended Practice. Electromagnetic

susceptibility procedures for vehicle components (except aircraft) - SAE Jill 3; 1984

June.

Vrooman, G. F. E. An indoor 60 Hz to 40 GHz facility for total vehicle EMC testing. SAE

technical paper series 831001, SAE passenger car meeting. Dearborn, MI; 1983 June 6^9.

Heirman, D. N. Automated immunity measurements. Proc. 6th Symp. Tech. Exhibition on

EMC; 1985 March; Zurich, Switzerland.

Crawford, M. L. Comparison of open-field, anechoic chamber and TEM cell

facilities/techniques for performing electromagnetic radiated emission measurements.

IEEE Int. Symp. on EMC; 1983 August; Arlington, VA. 13-418.

Hermeling, G. C. Method for immunity measurements on home entertainment products using

the 2 meter TEM cell. IEEE Int. Symp. on EMC; 1983 August; Arlington, VA. 574-580.

EIA Interim Standard No. 32 (proposed). Immunity of television receivers and video

cassette recorders (VCRs) to direct radiation from radio transmissions, 0.5 to 30 MHz.;

revised Sept. 6, 1984.

27
[32] Faught , A. N. , Jr.; Dowell, J. T. ; Scheps, R. D. Shielding material insertion loss

measurements using a dual TEM cell systems. IEEE Int. Symp. on EMC; 1983 August;

Arlington, VA. 286-290.

[33] Wilson, P. F. ; Ma, M. T. Shielding effectiveness measurements with a dual TEM cell.

IEEE Trans, on Electromag. Compat., EMC-27(3): 137-1^2; 1985 August.

[3U] Wilson, P. F. ; Chang, D. C; Ma, M. T. Excitation of a TEM cell by a vertical electric

hertzian dipole. Nat. Bur. Stand. (U.S.) Tech. Note 1073; 1981 March.

</i

us

03

COasc<

CO

u[ 5

CO 0)a

00 / S. £ a= •H / \ UJ Cb /r X 3 a

1 s

\

CO

\ a ^^

\

IS

CO

cc

1 —

a

1

* * * aa

UJ a a.

C/3 ^ Cd

CO

LUu

C3

3a.Z3

3OE < a

a. t-b.

28
2b

*-g-H

outer conductor

_Z

upper chamber

2w

inner conductor

lower chamber

T

/

1-5

side wall

2a

Figure 3.2 Cross section of a TEM cell.

1.0

J I I I L-L

10.0

a/b

Figure 3.3 Capacitance characteristics of a TEM cell.

29
0.1 0.2 0.3 0.4 0.5 0.6

w/a

0.7 0.8 0.9

Figure 3.4 Cutoff wavelengths of the first few higher order gapperturbed modes in a TEM cell.

30
0)3

C

(0g

1 .6

1 .4

!.0

0.8

0.6

THEORY

EXP

"'-0.5

0.6

-1.0 -0.8 -0.6 -0.4 -0.2 0.2 0.4 0.6 0.8 1.0

(a) b/a = 1, w/a = 0.83

x/a

0)

ex:

1 .5

1 .2 -

1 .0 -

0.3 -

0.6

-1.0 -0.8 -0.6 -0.4 -0.2 0.2 0.4 0.6 0.8 1.0 x/a

(b) b/a = 0.6, w/a = 0.72

Figure 3.5 Normalized magnitude of the electric field

of the TEM mode inside an RCTL.

31
FUtcr box L«adt covered with 2 inch

conductive tape.

Figure 3.6a EUT near floor of TEM cell for minimum

exposure of leads to test field.

FUter box Leeds covered with 2 inch

eoadurtive tape.

Figure 3.6b. EUT centered in test zone midway between

septum and floor.

32
ADDER

2 CHANNEL

XY R£COfU)£l PWH MTR

Pi--

SWEEP

GEN

HI PWR AKP

L0W-P4SS

FILTER

SIGNAL SAMPLER

COAXIAL

SWITCH

FREO COUNTER

PWR MTR TEST

MONI TOR

DUAL

DtRECTIONAL COUPLER

EUT • Eqwpmtnl Undtr Tt»t

30 n

LOAO

FiRure 3.7 Block diagram of system for susceptibility testing

of equipment at frequencies ^10 MHz.

Figure 3.8 Block diagram of automated TEM cell susceptibility

measurement system.

33
(a)

^TAPER ^RCTL

"TAPER "RC7L

Z Z RCTL T.\PER

(b) "Rc: Taper

Figure 3.9 (a) The equivalent T-network circuit for a small

scattering obstacle; and

(b) The TEM cell transmission-line circuit,

including the small obstacle loading.

0)u

c

c

6

3C

0)

l4-i o

<u "O3

4-1 •Hc

bC

ce S

60

58

56

54

52

50

48

46

44

42

40

"1 ' r

V. /

/ . \

\ * y

_L

t-*: MEASURED DFiTFI THEOPETrCFiL VRLUE2

J L.

50 100 150 200 250 300 350

Frequency (MHz)

400 450 500

Figure 3.10 Magnitude of the input impedance for an empty TEM

cell terminated with a 50-ohm load.

34
G

O

c

Q.e

3

p.C

4J

60

58

56

54

52

50 U—•-

48

46 -

*? 44

c

. 42

40

/ *

^

**** MERSURED DRTR

THEORETICRL VRLUES

50 100 150 200 250 300 350 400 450 500

Frequency (MHz)

Figure 3.11 Magnitude of the input impedance for the same TEM ceil

(as in fig. 3.10) loaded with a conducting sphere of

5.5 cm in diameter.

35
CO

1-1

4-1 u

o.

(U

Eo

1.12 -

1.09 -

1.06 -

1.03 -

1.00

0.97 -

0.9A -

0.91 -

0.88 _

0.85

1 '

\

'

\

' \

^

\ **** EY ++++ HX

OOGO EY/HX CWBVE IMFZDRNCE:?

+, V

,^ * * »., > I 1 fc.

NS.

' ^'»"'" 'j V Q u '^ »—i*fc «.

\ \

-2 -1.6 -1.2 -0.8 -0.4 0.4 0.8 1.2 1.6 2

z/a

Figure 3.12 Field perturbation in the same TEM cell (as in fig. 3.10)

by a conducting sphere of 5.5 cm in diameter at a frequency such that ka = 1.4

36
"lexAex

\\

"imx/^

mmy/'Ar^my

m

Z!! mz/ ^mz

Figure 3.13 An unknown electrically small source is considered to

be made of equivalent three orthogonal electric and

three orthogonal magnetic dipoles

TEM CELL

INSTRUMENTATION

FOR POWER AND

PHASE MEASUREMENT

COMPUTER SYSTEM

Figure 3.14 Emissions testing measurement system.

37
iY/y'

One half

TEM cell

Figure 3.15 Two EUT orientations in the TEM cell

38
y,z'

One half

TEM cell

y y'

(b)

Figure 3.16 Another two EUT orientations in the TEM cell

39
AY/X'

One half

TEM cell

(a)

(b)

Figure 3.17 Final two EUT orientations in the TEM cell
il

O

w ^ ^ w

u 3 e

Q. rf 3 cVE c "^ CL i 3 -1 u

Q. 0-

•D 0) O c

'4.' iH

C8 ^

•^

•O T3

cs ;^ ;h cq x: (M

;h

M «H •Hw -a

>. o

PH t, CB -H C 3

«8 O*

0) c ;h •H

esu E

+J •o «3

0) c E O

•H +J •M •!-> < U '«-'

«HE ^

(U ^

4J

O

to c; >.

(0 su M H co a E

tc

u c 3 -H m in a 3

Q)= B sE U E« ;^ (U h «H E

bfi CQ CO CO ^ C

•O «H

•H

X CO y CO o

-H c ^ E >»

CQ 0) CO

GO

cnu3

41
Table 3,1 Small-gap modal equations applicable to symmetric TEM cells.

Mode

[ref]

Modal Equation (f, = — -J (MHz) ) C 7T D

TEoi

[13]

TI b 2a -'- X tan X = - H [in (—) + R^^ ] 2 a Tig TE

01

R,. - I

I (cothM-1) cos^£j^)

'^01 p=l P

^ ^

T^IO

TT , bv

TEll

[13]

X = [y2 + (^)' ]l/2

cos2 (|i) ' "9 2a TE^^

2a

1 (2p+l)^b 2 2p+l

R = 2 I coth

—^

-1 cos (-^^ Tig TE p=l 2p+l 2a 2a

11

TE02 X = TT

TEi2,TM^2 X = . [1 + {|^)2]V2

TE20

b

TE2I

[13]

2cos2(-^) coty

cot X a ' ^ 2 ,a. ^ ,2a, ,.TTg^ ^

. -—— + = - (-) [zn(— -cos2(-^) + R ] X y TI b Ttg a TE

y = [x2 - (tt-)2 31/2

a

CO R^^ = I - (coth i— -1) cos2 (^ TE p=2 pa a 21

[13]

X = [y2 + (^)2 31/2 ;

^^"^ _ 2 ,a 2a 2 1 y T, ^^ ^^9^ ^ 'tm^^^

4a - ^ ^ (2p+l)^b ^^ ^ ,(2p+l)ng.

R = — I [coth 1] J [ 3 TM ^9 p=l 2a 1 2a

11

42
Table 3.2 Normalized field distribution in a TEM cell with a = 25 cm,

b = 25 cm, w = 20.6^4 cm; (a) x component and (b) y component.

y = b 25 0.000 0.000 0.000 0.000 000 0.000

20 0.000 0.060 0.129 0.208 278 0.307

15 0.000 0.108 0.245 0.422 600 0.680

10 0.000 0.127 0.311 0.620 1 029 1.237

5 0.000 0.090 0.248 0.647 1 684 2.285

0.000 0.000 0.000 0.000 000 3.603

10

(a)

15 20 25

y = b 25 0.824 0.793 0.698 0.530 0.289 0.000

20 0.853 0.825 0.736 0.568 0.315 0.000

15 0.935 0.917 0.852 0.699 0.410 0.000

10 1 .049 1.025 1.051 0.977 0.652 0.000

5 1 .153 1.186 1 .298 1.499 1.343 0.000

1 .196 1 .245 1.431 1.986 6.640 0.000

5 10 15 20 25

(b)

Table 3.3 Normalized field distribution in a TEM cell with a = 25- cm,

b = 15 cm, w = 18.025 cm; (a) x component and (b) y component

y = b

y = b

15 0.000 0.000 0.000 0.000 0.000 0.000

12 0.000 0.024 0.067 0.143 0.220 0.249

9 0.000 0.040 0.121 0.284 0.462 0.517

6 0.000 0.043 0.141 0.410 0.763 0.871

3 0.000 0.028 0.101 0.440 1 .247 1.112

0.000 0.000 0.000 0.000 1.969 1 .254

10 15 20

10 15 20

(b)

25

(a)

15 0.966 0.946 0.872 0.698 0.394 0.000

12 0.972 0.956 0.890 0.724 0.41

1

0.000

9 0.989 0.981 0.944 0.807 0.464 0.000

6 1 .010 1 .015 1 .028 0.979 0.557 0.000

3 1 .028 1 .046 1 .120 1 .311 0.645 0.000

1.035 1.058 1.164 1.664 0.000 0.000

25

43
Chapter ^. SHIELDING EFFECTIVENESS OF MATERIALS

How best to achieve adequate shielding is a concern of both the users and manufacturers

of electronic equipment. Shielding here refers to the practice of protecting some region from

electromagnetic (EM) radiation due to outside sources (susceptibility) or to the avoidance of

leaking unwanted radiation (emission). Shielding performance depends on both the electrical

properties of the materials used and on the construction of the shield (joints, seams,

apertures, geometry, etc.). This chapter is concerned with the basic problem of determining

material shielding properties. Traditionally, shielding depended on the use of metals whose

EM properties are well understood. Increasingly though, complex materials are replacing

metals and the intrinsic shielding effectiveness (SE) of these new materials may be difficult

to predict. As a result, there is significant interest in developing measurement techniques

which yield meaningful SE data. Two examples serve as illustrations.

Molded plastics have largely replaced metal boxes as housings for electronic equipment in

commercial applications. Plastic is basically transparent to EM radiation; therefore, some metal-like property must be added to insure proper shielding. Existing approaches include

conductive sprays, metal fibers injected during the molding process, foil inserts,

metallization, electro-plating, and others [1]. Shielding-associated costs may well exceed

] 0% of the production cost [2]. Thus, it is important to make the most cost-effective choice

among shielding methods, particularly when large production runs are involved. Recent FCC

regulations applicable to commercial electronics combined with the adverse consequence of

failing to pass FCC testing make the need for accurate SE data at the design stage all the more acute.

Avionics is another area where measured SE data are needed. Metal aircraft frames are being replaced partially or wholly with light-weight, high-strength composites and laminates.

These materials may or may not adequately shield the sensitive electronic equipment housed

inside an aircraft. As in commercial electronics, circuits continue to be miniaturized,

operating at ever lower power levels. As a result, the amount of spurious radiation necessary

to disrupt normal operation is also decreasing. Again, the need for reliable SE data at the

design stage can be critical.

Most SE test procedures essentially follow the arrangement depicted in Fig. M. 1 . Power

from a transmitting antenna ( P^ ) is coupled to a receiving antenna with no test material

present (P ) and then with the test material introduced (P'), as shown in Figs. 4.1a and b

respectively. The ratio of the received powers gives the resulting insertion loss ( IL)

,

IL = 10 log(P /P') dB (M.I)

Insertion loss represents a quantitative measure of SE provided by the test material. Thus,

the basic problem is quite simply stated. The difficulty is to separate the actual material

properties tested from external factors such as antenna types used, orientations, antenna

separation distance, field distribution, transient effects, etc. [31. Is one truly measuring

shielding when the material is inserted, or perhaps some other EM characteristics such as a change in an antenna input impedance? The present discussion will investigate these

considerations based on a fairly general theoretical model. The emphasis will be on the

importance of recognizing and understanding the effects of these factors when interpreting SE

data.

4.1 Theory -- Small-Aperture Coupling

The simplest model of an EM shield is an infinite plane sheet separating a source from a receiver. This case is fundamental to understanding shielding theory. Much work has been done on this problem including Schelkunof f

' s basic transmission line description [^l-ES] applicable

to planar-sheet, cylindrical, and spherical shielding, as well as the important case of lowimpedance (magnetic-field) shielding [9]-[12]. An analysis of the plane-shield case reveals

the basic mechanisms by which a material provides shielding: reflection, absorption, and

internal re-reflection [9]. Realistically, however, infinitely large test samples are not

available nor practical. A number of test procedures involve covering an aperture with a piece of shield material. Therefore, it is instructive to consider the problem of coupling

through a small aperture. Our intention is to use this model as a means of illustrating som$

of the variables affecting SE measurements.

44
Consider the geometry shown in Fig. M.2a. A set of fields, E. , H. , is incident on an empty aperture in a perfectly conducting screen. A cylindrical coordinate system (p, tf), z)

will be located at the aperture center, with the z axis directed normally to the aperture

surface as shown. If the aperture is electrically small, that is of dimension much less than

a wavelength, then the fields penetrating the aperture to the other side of the screen (z > 0)

will be similar to those produced by a set of equivalent dipole moments. The formal

derivation of small-aperture coupling is well detailed elsewhere [13]. The present discussion

requires only that we quote one basic result; namely, the fields produced by the equivalent

dipole moments are given by [14]:

ECR.o.) = -\—^

{( -^ - j -| ) [3R (R -Pq) - Pq]

R^ R k^ . . k k^ . - -^ [R X (R X Pq)] + ^ (J -| - p^ ) (R X Mq)}

(4.2)

-jk R k . . _

H(R,ai) = -T-^ f( 4 ^ J -^ )t3R (R • M^) - M^]

4TT r3 r2

k^ . . k k^ . - -^ [R X (R X Mq)] - c(j -| - -^ )(R X Pq))

R

where a time convention of exp(jwt) has been assumed, k_ = co/c, c is the speed of light, R =

2 2 1/2

(p + z ) is the distance from the aperture center to the observation point, e^ is the

free-space permittivity, R is the normalized position vector, [ R = (pp + zz)/R ], P^ is the

equivalent electric dipole moment due to the aperture fields, and M is the equivalent

magnetic dipole moment.

The dipole moments depend on 1) the incident fields exciting the aperture and 2) the

geometry of the aperture. This situation leads to the following standard decomposition:

P_ = S • e^ECO) = z e^ a E (0)

e e z (4.3)

M-Cm= a„- H(0) = p a H (0) m p

where a and a are the electric and magnetic aperture polarizabilities dyadics (the geometry

term), and E(0) and H(0) are the incident fields at the aperture center 0. The electric

dipole moment is normal to the aperture (a = z a z) while the magnetic dipole moment is

tangential (a = p a p) where, as implied by (4.3), we have oriented the p axis to coincide

with the tangential component of the incident magnetic field.

Equations (4.2) and (4.3) together yield expressions for the penetrating fields. For the

near-field (k R << 1) and far-field (k R >> 1) special cases, we find respectively that:

a E (0) -jk-R . - - o 2 rfu ^ e z e

-' r 3pz ^ ^ ,3z ,.-1 E(R,a)) = J- 5— [p -^ + z (—X - 1)J

R^ R R (near-field)

45
and

H(R.-) =-^a H (0) -jk-R . , 2 . . ^ nL„^_ ^0 [p( 3|_ _ T) , , 3pz

^ R^ R R

(U.il)

ECR.co) = - ^

^ e-J'V r P

[ a (p I

_ , L) ,^ E^(0)

^ol%"p^°)J

(far-field)

HCR.o)) = - 17^ ^0 e-JV >^P|-^l) %"p^°^

-1 ^ a E (0)

Iq R e z

where rir, = 12077 is the free-space impedance, and $ is the unit vector,

A typical receiving antenna will respond primarily to either the electric or magnetic

field with the received power proportional to the square of the appropriate component in the

far-field case. Thus, each of the far-field components in (H.k) leads to a type of insertion

loss measurement via (4,1). Idealizing the problem, we let

antenna's response to any of the field components, denoted by F(R,co)

loss should then behave like

A represent the receiving

The far-field insertion

IL = 20 log

F(R, ai)'A

F'(R,a))-A

(4.5)

where the prime denotes the loaded aperture case, as depicted in Fig. 4.2b. Although (4.5) is

strictly true only in the far-field case, we shall also apply it to near-field expressions in

order to demonstrate the types of variations possible.

Equation (4.5), with one of the field components in (4.4) inserted, may now be used to

examine some of the factors influencing insertion loss measurements. It should again be

emphasized that (4.5) does not represent any specific measurement procedure per se; rather, it serves to illustrate some basic properties likely to be encountered in shielding studies

featuring coupling through an aperture. We will consider the following variables: receiving

antenna orientation, incident field distribution, and shielding material parameters.

4.2 Effects of the Receiving Antenna Orientation

Consider Fig. 4.2a. Intuitively one would probably locate the receiving antenna along the

z axis, or boresight. In this case, p = 0, z * 0, and (4.5) yields

IL = 20 loj a E (0) e z

a' E' (0)

e z

IL,

(z • A * 0, E-measured, near field),

IL = 20 lof a H (0) m p a' H'(0)

m p

IL, (4.6)

(p • A * 0, H-measured, near field)

46
(() • A * 0, E-measured, far field

H ^ p • A * 0, H-measured, far field ^

where, as before, a prime denotes a loaded aperture quantity. In the near-field case,

insertion loss depends on either electric-field coupling (IL ) or magnetic-field coupling

(IL„) through the material. If, however, the receiving antenna is located in the far field,

H (4.6) reveals that we are considering IL regardless of the type of antenna used. For a

ri typical conductive shield material, the difference between XL and IL can be dramatic. Thus,

h n an insertion loss measurement can change significantly as we move from the near field to far

field, as would be expected.

Another consideration is the effect of loading the aperture on the incident waveform.

Ideally, the incident fields would be unchanged, that is E (0) = E'(0) and H (0) = H'(0). If

z z p p this is indeed the case, then

ILg -> 20 log |ag/a;|

IL„ -> 20 log la /a' I H 'mm'

(M.7)

a desirable result, since we are primarily interested in measuring the material shielding

properties which are contained in a' and a'. In general, the loading of the aperture could

affect the transmitting antenna's characteristics, especially if the aperture is loaded in the near field of the transmitter. In fact, if the aperture is viewed as an impedance, loading

the aperture might actually provide a better impedance match leading to greater field

penetration with the shield material in place. This type of "gain" behavior has been observed

in practice.

Suppose we now shift the receiving antenna from a boresight location to a grazing

orientation (z = 0, p * 0). Insertion loss should then behave as follows:

IL„ (z • A * 0, E-measured, near or far field),

Ei

IL„ (p • A * 0, H-measured, near field) (4.8)

ri

IL„ ((}) • A ji 0, H-measured, far field)

bi

We now find a different set of field coupling dependencies.

For receiving antenna orientations between these two extremes, both electric- and

magnetic-field couplings will contribute. The implication of the above discussion is

apparent; the choice of receiving antenna location can significantly affect an insertion-loss

measurement, independent of the particular shield material being tested. One needs to

recognize what type of coupling is being measured if results are to be given a proper

interpretation.

4.3 Effects of Incident Field Distribution

Three types of incident field are usually associated with shielding measurements: plane

wave, high impedance, and low impedance. The choice of incident-field type can certainly

affect an insertion-loss measurement, as can be demonstrated via (4.5).

For a plane wave with an assumed normal incidence, both the electric and magnetic fields

will be tangential to the aperture. Thus, E (0) = 0, and in all cases (4.5) reduces to

IL -> ILu (normal plane wave) (4.9)

H

47
For a plane wave incident from a grazing angle, two polarizations are possible: E in or perpendicular to the plane of incidence. In the latter case, both E (0) and H (0) are zero

z p and negligible fields will be coupled through the aperture. For E parallel to the plane of

incidence, both E (0) and H (0) are present with E (0) = n^ H (0). If the receiving antenna

z p z p is also in the far field, we find that,

IL = 20 log

E,(0) If [p f

- • A

e;(o) If

:p- f

-

^ ff^ "•; ^ 1

o;i • A

(E-measured, far field, grazing plane wave)

i^.^0)

IL = 20 log

E,(0) if [P I

- z |] a^ let ] R e' E:(0) ll [p I

- z |] a- l«'l R e'

(H-measured, far field, grazing plane wave).

If the receiving antenna is located in the near field, then IL ^ IL if E is measured, while

hj IL * IL„ if H is measured. Equation (M.10) indicates that both the material electric and

n magnetic shielding properties are important for the grazing plane-wave case, whereas a normal

plane wave (M.9) largely tests the material magnetic shielding properties only. As discussed

in the previous section, the effects of a and a in (U.10) may be decoupled individually by

reorienting the receiving antenna.

If a high-impedance incident field is applied, as produced by an electric dipole near the

aperture for example, then H (0) << E (0) and

P z

IL -* IL„ (high-impedance field)

h C^.ll)

independent of the receiving antenna type. Analogously, a low-impedance field implies that

H (0) >> E (0), thus

P z

IL -* IL„ (low-impedance field)

H (4.12)

in all cases. Comparing (M.12) and (4.9), we see that insertion-loss measurements using

coaxial magnetic loops should yield results similar to those based on a normally incident

plane wave if aperture coupling is used. If we were coupling fields through a plane sheet of

material, this conclusion would not hold since in one case the wave impedance is low and in

the other (plane wave) the wave impedance is n . Again, it should be emphasized that the range

of values defined by IL„ and IL„ can be large; thus the choice of incident field significantly

h H affects the resulting insertion-loss measurements. This point can be explained by considering

some realistic material parameters.

4.4 Material Factors

The loaded aperture polar

zabil ities are difficult to specify in general, as is the

effect of loading on the incident field. If, however, we limit our attention to a circular

aperture of radius r and retain only the first-order terms in l<p,r , then approximate loaded

aperture polari zabi 1 ities are available for the thin conductive sheet model depicted in Fig.

4.2b, based on a formulation by Casey [15]. We find that

48
o j 160 k^r oh „ „

a 1 + 2iT onR m c

and

a

e

-> 00 a'e

where o and h are the material conductivity and thickness, and R is a possible contact

resistance between the material and the aperture plane conductor. As indicated, to this order

of approximation, a' is negligible. Note that the above magnetic polarizability ratio depends on material properties (o and h) as desired, but also on the aperture size (k„r) and the

contact resistance R which have no intrinsic relationship to the material EM shielding

properties. Given any SE measurement procedure, it is important to recognize these extrinsic

factors even if they cannot be isolated out.

Equation (4.13) may be inserted into (4.6) to generate some representative values for

IL„. We see that IL^ «> for this approximation. Clearly, this will not be the case in

H b practice, but it is well known that any reasonable conductor will provide high levels of

shielding against electric fields. Magnetic-field shielding, as measured by IL^., tends to be

H

7 more of a problem. For example, suppose an aluminum sheet [o = 3-72(10) S/m] of thickness h

-4

= 1.27 (10) m [5 mils] is deposited on a plastic substrate. This might model a conductive

spray applied to the plastic housing of some piece of electronic equipment. Assuming that

H (0) = H'(0), and choosing k^r = 0.2 as a typical value, we obtain

P P

IL^ = 10 log 1.51(10)^ n2

1 + 2.96(10) R

c

dB. (4.U)

If the contact resistance is zero, a value XL = 103.6 dB results. This would indicate quite

H good magnetic-field shielding. However, suppose some contact resistance exists, say R =0.01

fi. Then IL„ reduces to 54.1 dB. If R rises to 1 fj, IL,, = 14.3 dB. Thus, a ]-Q resistance

n C H reduces IL„ by 89.3 dB, a rather significant change which has nothing to do with the material

H itself. In fact, variations in R could mask actual shielding properties. Contact resistance

could be a problem where conductivity-filled plastics are used, since surface resins could

cause R^ to be relatively high. The above example also indicates that the range of IL„ to IL„

C h n could be large. As discussed, changing the antenna orientation or incident field can shift IL between these values. One is tempted to speculate that given any test material, changing

these various factors could yield any level of insertion loss desired.

Insertion-loss measurements are a valuable tool to the EMC engineer; however, this tool

must be used carefully. The basic point is that insertion-loss data may well depend more on

the measurement procedure than on the material tested. Thus, when one attempts to evaluate

the shielding capability of various materials, it is very important to understand how

shielding data are obtained.

4.5 Measurement Techniques

We have seen that the key to understanding SE data is to evaluate the measurement

procedure itself. We now turn our attention to the practical task of obtaining shielding

data, reviewing specific methods which have been tested at NBS.

The particular methods of measurement studied at NBS are: a variation of MIL-STD 285

[16], a pair of circular coaxial transmission-line holders, a time-domain system in free space

and through an aperture in a shielded room [17], a dual TEM cell [18], [19], and an apertured

TEM cell in a reverberating chamber [20]. As expected from the above discussions, there can

49
be substantial disagreement among the measured values of SE for the same material when these

different methods are used.

A longstanding SE test procedure is the use of a shielded room approach based on a modification of MIL-STD 285 [16]. A source is placed inside a shielded room, a receiver

outside, and coupling is via an aperture in one of the walls. This method is simple, and

shielded rooms are widely available. However, repositioning the source antenna can vary

results up to ±40 dB because of room resonances [21]. In view of this wide variation of

results, only by taking a tremendous number of measurements and "averaging" can any meaningful

data be obtained. This is prohibitively time consuming and costly. Reproducing results using

different shielded rooms is extremely difficult. The data shown in Fig. ^i . 3 are for a piece

of plastic-aluminum-plastic, layered, fabric-like material obtained in a shielded room of

2 dimensions 2.34m X 2.10m X 3-Om and with a 46 cm aperture. Tuned dipole, biconical, or TEM

horn antennas were used at two different distances and two sets of data were taken with and

without material over the aperture. Data were taken at close frequency intervals in order to see effects of mode resonances. Due to the large number of modes that can exist within the

shielded room, the orientation and impedance of the fields exciting the aperture is largely

unknown. Therefore, it is difficult to specify whether shielded-room measurements simulate

near-field or far-field shielding conditions.

True far-field testing using an infinite sheet of the material under test (MUT) is

impractical. A realistic alternative is to use a waveguide to excite and receive the fields

resulting in a simple, well isolated system. If the waveguide is to simulate a free-space

plane wave, then a two-conductor transmission line supporting a transverse electromagnetic

(TEM) wave is the logical choice. Two such coaxial holders will be considered. The most

commonly used holder at present is that proposed by the American Society for Tests and

Materials (ASTM) Committee D09.12.14 (for EMI shielding) as one of the emergency standards (ES

7-83). The holder is essentially an expanded section of 50-n circular coaxial line which may

be disassembled to allow the insertion of an annular (washer shaped) test sample [22], as shown in Fig. 4.4. This fixture features continuous inner and outer conductors of dimensions

4.35 cm and 9.90 cm respectively.

A coaxial line such as the ASTM holder may be analyzed as a transmission line with the

MUT represented as a loaded section [20]. However, if we recognize that the coaxial holder is

primarily a low-frequency device, it should be adequate to model it as a simple circuit as shown in Fig. 4.5, where Z is the characteristic impedance of the transmission line (50 Q)

and Z. represents the impedance of the MUT. Based on this model IL should behave like

Z IL = 20 |1 + -^ I . (4.15)

The primary difficulty is that contact impedance Z between the MUT and the conductors can degrade results. Because the conductors are continuous, the contact impedance will appear in

series with the MUT [23]. Thus, in (4.15) we would replace Z by Z + Z , or

Ij Li

Z IL = 20 log|l ^ ^(^ : ^ )

| . (4.16)

L) U

This expression indicates that contact impedance tends to degrade the measured insertion loss

since the effective load impedance (denominator) is increased.

A second type of circular coaxial holder is a flanged version developed at NBS [24], as depicted in Fig. 4.6. It is similar to the ASTM design, except that two large flanges are used to hold a round disk of the MUT in place and capacitive loading couples the TEM mode

through the MUT. It has an inner conductor diameter of 3.2 cm and an outer conductor diameter

of 7.6 cm. The flange has an outer diameter of 13.3 cm. The "unloaded" reference measurement

involves two pieces of the MUT which match the dimensions of the flanges while leaving the

space between the inner and outer conductors empty. This tends to overcome the contact

impedance problem except at frequencies too low to generate sufficient displacement currents

in the MUT. A circuit model for the flanged holder is more complicated due to the capacitive

effects [20]. However, if the capacitive coupling is strong, then analysis predicts that IL behavior will be similar to that of the idealized circuit shown in Fig. 4.5. In fact, we need

50
good conductivity on only one side of the MUT to reduce the flanged holder XL expression to

that of equation (4.15).

Figure 4.7 shows measured XL data up to 1 GHz from both holders for an aluminum-Mylar

layered material. At these frequencies the aluminum-Mylar sample is electrically thin, thus

we should have [20]

Zq/Z^ - ripOd , (4.17)

where n,^ is the free-space wave impedance (377 fi), a is the conductivity of the MUT, and d is

its thickness. Inserting this into (4.15) we see that an electrically thin MUT in a coaxial

holder should ideally yield

XL = 20 log|l + J

npOdl . (4.18)

This expression is independent of frequency and represents reflection alone. The flanged

holder yields a flat XL curve of approximately 24 decibels. The ASTM holder yields a significantly lower XL level even when the material is silver painted to lower the contact

impedance. Xf od is known for a particular thin material, then (4,18) should allow one to

predict the expected XL level. Conversely, measured results (when flat) yield an effective ad

value; in this case XL = 24 decibels implies that od = .079 for the aluminum-Mylar material.

This value will be used later in discussing the near-field techniques.

Figure 4.8 shows data for the plastic-aluminum-plastic material. Because the surface of

this material is insulating, we expect that contact impedance could be a problem. Again the

flanged holder yields a flat XL of approximately 28 decibels, while the continuous conductor

configuration data are more variable, even when the MUT is silver painted.

The upper frequency limit for usage of these coaxial holders is approximately 1 GHz while

the lower limit (here 1 MHz) is determined largely by equipment. The dynamic range for these

holders is 90-100 decibels and 15-30 minutes are required to generate a typical XL curve.

The coaxial holders are limited in frequency due to the appearance of higher-order modes

which perturb the desired TEM mode field distribution. In order to gain plane-wave SE data at

higher frequencies a time-domain signal approach is being investigated. The MUT is either a large sheet, as shown in Fig. 4.9, or a small sample mounted over an aperture in a large

conducting screen (or shielded room). A short pulse is used as the source signal. Unwanted

signal paths to the receiving antenna can be time windowed and only the direct path signal

retained. This makes the large sheet, or the conducting screen, appear to be infinite in

extent for some short interval. Time-domain data may be transformed to the frequency domain

using Fourier transform techniques. Xf the source antenna is X/2it removed from the MUT, the

arriving fields should be approximately plane wave. Coupling may be modeled in the frequency

domain as a sequence of transmission functions representing propagation to the MUT from the source antenna (P.), transmission through the MUT (T), aperture transmission (A) if

applicable, and propagation to the receiving antenna (Pp) [20], [25]. For an insertion-loss

measurement the propagation functions divide out and we are left with

IL = 20 log|l/T| . (4.19)

For a highly conductive screen excited by a normal plane wave we find [25]

2n

n sinhTd ' (4.20)

1/2 1/2 where n = (jwy/o) is the material intrinsic impedance, and Y = (jtoyo) is the

propagation constant in the material. If |Yd| is small which is the case for an electrically

thin MUT, then (4.20) reduces to

T = 2/nQod . (4.21)

If this expression is inserted into (4.19), we see that IL is approximately given by

51
IL = 20 log|^ riQodl . (M.22)

This agrees with the coaxial holder expression (4.18) if n^od >> 1.

Figure ^1.10 shows IL data for the same layered aluminum-Mylar material, obtained with the

time-domain technique. Both a large sheet and an apertured screen are used. In the latter

case the MUT may be mounted either on the source or receiver side. The IL data due to large

sheet free-space simulation tend to be around 28-30 decibels which is somewhat higher than

found in the coaxial holders. For the MUT mounted on the source side over a screen aperture

we see basic agreement to the large sheet data above 1 GHz. The aperture significantly

reduces coupling below 1 GHz thus the dynamic range there is poor. If the MUT is mounted on

the receiving antenna side of the screen, results are further affected. The fields exciting

the MUT are no longer plane wave, rather they are the aperture fields. Thus we now see resonance behavior. The implication is that if small samples are to be used over apertures,

they should be mounted on the source antenna side if far-field simulation data are desired.

Figure 4.11 shows data for the plastic-aluininum-plastic sample. Again the large-sheet IL data are somewhat higher than in the flanged coaxial holder (30-34 dB vs. 28 dB) , but overall

agreement is quite reasonable. The source-side data agree well with the large-sheet results

except below 1 GHz, where as noted above, dynamic range begins to fall.

The frequency range for this method is from 200 MHz to 3.5 GHz. Shorter pulses would

allow the upper frequency limit to be extended even further. Measurement time is fast (5-10

minutes) and repeatability is good. The present dynamic range is 50-60 decibels if a large

sheet of MUT is used, and is lower for an aperture measurement particularly below 1 GHz.

In addition to far-field shielding data, it is also important to test the material nearfield shielding performance. One such near-field method is the dual TEM cell (DTC). A single

TEM cell is a section of expanded rectangular coaxial transmission line, as discussed in

Chapter 3- The dual TEM cell fixture. Fig. 4.12, uses one cell to drive another via an aperture in a common wall. Coupling can be well modeled using small aperture theory. The

sensing cell has two output ports. By summing (Z) and differencing (A) these two output

signals, the coupling of the normal electric-field (high-impedance) and tangential magneticfield ( low- impedance ) components through the MUT may be monitored separately and

simultaneously. One may show that [20]

IL(I) = 20 logla /a'

' e e

and (4.23)

IL(A) = 20 logla /a' I,

' m m'

where a is the electric polarizability in the direction normal to the aperture, a is the

e

t^ J ^ ' m magnetic polarizability tangential to the aperture and normal to the direction of propagation

in the TEM cells, and the prime denotes the presence of the MUT as before. Thus IL(I) gives a measure of the electric-field penetration through the MUT while IL(A) measures the magneticfield penetration. It should be noted that the aperture itself also influences these

quantities.

Sample results for the aluminum-Mylar material are given in Fig. 4.13. As expected of a good conductor, IL is greater for the electric-field component (sum power curve) than for the

magnetic (difference power curve). A theoretical curve for magnetic-field IL is also given

based on loaded aperture polarizabilities derived from Casey [15], [20] and the coaxial holder

od value discussed earlier. Agreement is quite good. Resonances associated with higher-order

modes in the TEM cells begin to appear around 770 MHz; thus practical use of the present

fixture is limited to frequencies below 1 GHz. These data may be compared to the 24-30

decibels IL characteristic of the two far-field simulation techniques. The magnetic-field IL curve is expected to be less than the plane-wave level, approaches zero along with the

frequency, and is tending toward the free-space simulation level with increased frequency.

However, the electric-field curve does not always exceed the free-space level as expected.

Figure 4.14 shows sum and difference IL curves for the plastic-aluminum-plastic MUT. Again

electric-field shielding exceeds magnetic, a distinct resonance appears around 770 MHz, and

the theoretical difference-power curve is in good agreement.

52
The dynamic range of this system is 50-60 decibels, repeatability is good, and

approximately 30 minutes are required to generate a single curve.

An alternative to the DTC is to use a reverberation chamber to excite an apertured TEM

cell. A reverberating chamber is a modified shielded room which allows us to generate a highfrequency, statistically known field. (Details on reverberating chambers are referred to

Chapter 8.) The analysis is similar to that for the DTC [20]. Figure 4.15 shows IL data for

the aluminum-Mylar sample. Results are very similar to those obtained in the DTC except that

the electric-field shielding (sum power) tends to be somewhat lower and more variable. Half

of the DTC was used for this study; thus the upper frequency limit is the same (less than 1 GHz). The use of smaller cells will enable us to extend coverage to higher frequencies.

Figure 4.16 shows data for the plastic-aluminum-plastic material. Proper use of the

reverberating chamber requires that it be highly multi-moded. The dimensions of the NBS

chamber are such that below 200 MHz the number of modes is insufficient; thus in both curves

we see poor results below this frequency. The dynamic range of this system is 90-100

decibels, and repeatability is good. However, the time required to produce a single curve is

long, in the order of 3 to 4 hours.

Some other measurement methods have also been tried by the EMC community. For example,

near-field testing using a dual box has been suggested by ASTM as another emergency standard

[26]. This technique is essentially a scaled version of a shielded room and therefore suffers

the same reproducibility problem. Conductivity measurements via the transfer-impedance method

[27] can be used to predict plane-wave shielding. Additional variations of the coaxial line

and dual box methods have also been reported [28].

REFERENCES

[1] Miller, M. B. Materials and techniques used in EMI/RFI shielding of plastic chassis

for electronic systems. EMI/RFI Shielding Plastics Symposium Record (Society of

Plastics Engineering); 1984 June; Chicago, IL. 139-150;

[2] Armstrong, H. W. The long term environmental effects upon conductive coatings for

plastic enclosures. MIDCON/82 Conference Record, 28/2/1-5; 1982 November; Dallas, TX.

[3] Franceschetti , G. ; Papas, C. H. Steady state and transient electromagnetic coupling

through slabs. IEEE Trans. Antennas and Propag. AP-27(5): 590-596; 1970 September.

[4] Schelkunoff, S. A. Electromagnetic Waves. New York, NY: Van Nostrand; 1943. 303-315.

[5] Cowdell, R. B. New dimensions in shielding. IEEE Trans. Electromag. Compat . EMC10(1) : 158-167; 1968 March.

[6] Stirrat, W. A. USAECOM contributions to shielding theory. IEEE Trans. Electromag.

Compat. EMC-IOd): 63-66; 1968 March.

[7] Schulz, R. B. ; Plantz, V. C; Brush, D. R. Low-frequency shielding resonance. IEEE

Trans. Electromag. Compat. EMC-IO(I): 7-15; 1968 March.

[8] Babcock, L. F. Computer determination of electromagnetic wave shielding effectiveness.

IEEE Trans. Electromag. Compat . EMC-1 0( 1 ) : 331-334; 1968 March.

[9] Levy, S. Electromagnetic shielding effect of an infinite plane conducting sheet placed

between circular coaxial coils. Proc. IRE, vol. 24(6): 923-941; 1936 June.

[10] Moser, J. R. Low-frequency shielding of a circular loop electromagnetic field source. IEEE Trans. Electromag. Compat. EMC-9(1): 6-I8; 1967 March.

[11] Bannister, P. R. New theoretical expressions for predicting shielding effectiveness

for the plane shield case. IEEE Trans. Electromag. Compat. EMC-IO(I): 2-7; 1968 March.

[12] Bannister, P. R. Further notes for predicting shielding effectiveness for the plane

shield case. IEEE Trans. Electromag. Compat. EMC-1 1(2): 50-53; 1969 May.

[13] Collin, R. E. Field Theory of Guided Waves, New York, NY: McGraw-Hill; I960; 288-298.

53
[14] Taylor, C. D. Electromagnetic pulse penetration through small apertures. IEEE Trans.

Electromag. Compat . EMC-15(1): 17-26; 1973 February.

[15] Casey, K. F. Low-frequency electromagnetic penetration of loaded apertures. IEEE

Trans. Electromag. Compat. EMC -23(4): 367-377; 1981 November.

[16] MIL-STD 285 Attenuation measurements for enclosures, electromagnetic shielding, for

electronic test purposes, method of. DoD; 1955 June.

[17] Ondrejka, A. R. ; Adams, J. W. Shielding effectiveness measurement techniques. IEEE

Natl. Symp. on EMC; April 1984; San Antonio, TX.

[18] Wilson, P. P.; Ma, M. T. Shielding effectiveness measurements with a dual TEM cell.

IEEE Trans. Electromagn. Compat. EMC-27(3): 137-142; 1985 August.

[19] Faught , A. N. Jr.; Dowell, J. T. ; Scheps, R. D. Shielding material insertion loss

measurement using a dual TEM cell system. IEEE Int. Symp. on EMC; 1983 August;

Arlington, VA. .

[20] Wilson, P. F. ; Ma, M. T. A study of techniques for measuring the electromagnetic

shielding effectiveness of materials. Nat. Bur. Stand. (U.S.) Tech. Note 1095;

1986 May.

[21] Free, W. R. Radiated EMI shielded enclosures. IEEE Int. Symp. on EMC; 1967 July;

Washington, D. C. 43-53-

[22] Simon, R. M. ; Stutz, D. Test methods for shielding materials. EMC Tech., vol. 2; 39-

48; 1983.

[23] Week, R. A. Thin-film shielding for microcircuit applications and a useful laboratory

tool for plane-wave shielding evaluations. IEEE Trans. Electromagn. Compat. EMC-10(2):

105-112; 1968 March.

[24] Adams, J. V/ . ; Vanzura, E. J. Shielding effectiveness measurements of plastics. Nat.

Bur. Stand. (U.S.) Int. Rept . 85-3035; 1986 January.

[25] Ari, N. ; Hansen, D.; Schar , H. Electromagnetic pulse (EMP) penetration through

circular apertures in the frequency domain. Proc. IEEE, vol. 73(2): 368-369; 1985

February.

[26] Oberholtzer, L. C; Mauriello, A. J.; Stutz, D. E. A treatise of the new ASTM EMI

shielding standard. ITEM 1984; 174-178.

[27] Dike, G; Wallenburg, R. ; Birkin, J. Electromagnetic relationships between shielding

effectiveness and transfer impedance. IEEE Int. Symp. on EMC; 1979 October; San Diego,

CA. 133-138.

[28] Hariya, E. ; Masahiro, U. Instruments for measuring the electromagnetic shielding

effectiveness. IEEE Int. Symp. on EMC; October 1984; Tokyo, Japan. 800-805.

54
(a) Unloaded coupling.

P. P'.

(b) Loaded coupling.

IL = 10 log

Figure 4.1 A typical arrangement for measuring shielding

effectiveness of materials.

55
Aoerture

(a)

^- h

Aperture

-Test material

(b)

Figure 4.2 An arrangement for aperture SE measurement;

(a) Fields incident on an empty aperture,

(b) Fields incident on a loaded aperture.

56
N

N

N s § g m m o CN <r o m CM 00 o O o u •u 4-) o in o in in 00

r-\ rH r- 1 1 1 ON

r. ^

•>

N

N

N § g m O O <r o Csl CN v£)

i-H

O o O 4-1 4-1

4-1 in o o in o un rH <r 1 CM 1 m 1 CO

r. ^ .

•<

N

NN

N

g § §

in o o o CN o o CN en vX) 00

I-H

O o o o 4-1 4-) 4-) 4-1 o o o O in o 00

in rH o- r^ 1 1 1 1 o + 2 S ? 8 —»—

CSO) 3S s

t*

1^

CD

CD

in

.. to

in

4- CD ocD

cr 8

.9

<u

•HT)

0)

}-i 3

0}

CO

(U6

CO

COo

•H

4-1 CO

CO rHaI g CO C O 3 CO

<-{

U

CO

O I -H U

•H T3

4-1

C

CO CO

CO iH CO

CI. COc

CO

C

t-i 4J o c

M-i CO

CO 01

+-I I-H CO 0~ T3 -H

4-1 CO iH

CO 3 O EC

•H

CO3eoo

CO

<uu3

bO

•H

57
Figure 4.4 ASTM model of circular coaxial transmissionline holder.

Figure 4.5 Idealized coaxial holder circuit model.

58
13.3 cm

REFERENCE LOnD

Figure 4.6 The NBS flanged coaxial transmission-line holder,

59
ffi TJ

ino

_l2

O

u

z

50

40

30

20

-r T T

MERSURED IN: »»»»» NBS HOLDER

^-^ —^ RSTM HOLDER EDGES UNPRINTEn

+++++ RSTM EDGES SILVER PRINTED

IW» llll»MllBWl<ll'l»»l»Mir •——<• » *"

200 '400 E00

FREQUENCY (MHz

)

B00

-w 1

1 000

Figure 4.7 Insertion losses for an aluminum-Mylar sample measured

in the ASTM and flanged circular coaxial holders.

m

•o

m

o

_izo

UJ

Ul2

50

40

30 -

T T -r -r MERSURED IN: ««««» NBS HOLDER

.^ RSTM HOLDER EDGES UNPRINTt.D

+ -•- + + + RSTM EDGES SILVER PRINTED

200 400

FREQUENCY

600

(MHz ) B00 1000

Figure 4.8 Insertion losses for a plastic-aluminum-plastic sample

measured in the ASTM and flanged circular coaxial holders,

60
Material

Under Test

Computer

Impulse

Generator

Sampling

Oscilloscope

Figure 4.9 The time-domain measurement system,

m

T3

0)o

_Izo

u

I/) 2

60

50

40

30

20

10

-1—I—I—I—

[—1—I—I—I—I—I—I—I—I—I—I—I—I—I—

I—I—I—I—I

\—I—I—I—

»»» SHEET

+++ RPERTURE - MUT SOURCE SIDE

000 RPERTURE - MUT RECEIVE SIDE

-J—

I I I I. _i I I I I I I I i_ 500 1000 1500 2000

FREQUENCY (MHz)

2500 3000

Figure 4.10 Insertion losses for an aluminum-Mylar sample

measured with a time-domain signal.

61
IS

•0

If) oJ

zoM

I-U

1/1 z

60 .

50

40

30

20

10

r—I—I—I—r-

-r—I—I—I—I—I—I—I—I—I—I—I—I—I—I—I—I—I '

I

I I "^ >"

Hi M >

«** SHEET

+++ RPERTURE - MUT SOURCE SIDE

I I I 1 1 I I I I—I—1—I—I—I—I—I—I—I—I—

500 1000 1500 2000 2500 3000

FREQUENCY (MHz)

Figure 4.11 Insertion losses for a plastic-aluminum-plastic sample

measured with a time-domain signal.

kd-i

L i APERTURE

TOP VIEW

No 2

Nol

SIDE 7IEV END VIEW

Figure 4.12 A dual TEM cell. The particular model used at NBS has the

following dimensions: a = 9 cm, b = 6 cm, g = 2.2 cm, and

d = 5.08 cm, where d is the aperture side length, and a, b,

and g have the same meanings as those in Chapter 3 for a single TEM cell.

62
50

«««»«« SUM POWER IL

• /'••/' DIFFERENCE POWER IL

ooooo DIFFERENCE POWER IL. THEORY

200 400 600

FREQUENCY (MH2)

1000

Figure 4.13 Insertion losses for an aluminum-Mylar sample measured

in the dual TEM cell shown in Figure 4.12,

G0

50 - •«»«» SUM POWER IL

//yyy DIFFERENCE POWER IL

ooooo DIFFERENCE POWER IL. THEORY

400

FREQUENCY

600 1000

Figure 4.14 Insertion losses for a plastic-aluminum-plastic sample

measured in the dual TEM cell shown in Figure 4.12.

63
50

A5 -

w

ino

z

o

bi

V)z

J

,

1 1 \

, sss SUM POWER IL

ddd DIFFERENCE POWER IL

DIFFERENCE POWER IL, THEORY

2BB

FREQUENCY (MHz)

600 1000

Figure 4.15 Insertion losses for an aluminuni-Mylar sample measured with

an apertured TEM cell in a reverberating chamber.

n

ino

-Izo

u

inz

60

55

1

'

1 s ss ddd

'

1 SUM POWER

DIFFERENCE

IL

POWER

1

IL

1

SPI ^

DIFFERENCE POWER IL. THEORY.

45

40 . -

35 l\ A, ^^ -

30

/

S^-^.^^ /

^V^.^

25 / "V -^-^^

20 / A^^^y^:-:0{_ "-

15 - '' '.^^ -

10 ^

-

5 - ^/^ -

z-'^'^, 1 1 1 1

1

1 200 400 600

FREQUENCY (MHz)

BO0 1000

Figure 4.16 Insertion losses for a plastic-aluminum-plastic sample

measured with an apertured TEM cell in a reverberating

chamber

.

64
Chapter 5. ANECHOIC CHAMBER MEASUREMENTS

Microwave anechoic chambers are currently in use for a variety of indoor antenna and

EMI/EMC measurements. The prime requirement is that an appropriate transmitting antenna at one location within the chamber generates a plane-wave field throughout another volume of the

chamber of dimensions sufficient to perform tests. This volume is frequently referred to as a quiet zone and its "quietness", or reflectivity level, will determine the performance of an anechoic chamber.

The anechoic chamber at NBS is shown in a side view in Fig. 5.1. Pyramidal horns or open-ended waveguide (OEG) antennas are used as sources of chamber illumination at a position

in the access doorway with their apertures inside the plane of the absorber points on the

chamber wall. A cart on precision tracks located under the measurement axis can be moved

horizontally through a distance of 5 m by a stepping-motor drive system. There are gaps in

the absorber on the floor to accommodate each rail.

The purpose of this chapter is to discuss how to establish a standard EM field, and the

methodology for evaluating the corresponding errors associated with antenna and EMI/EMC

measurements in an anechoic chamber.

5.1 Measurement of the Net Power to Source Antennas

One potential error in the standard EM field to be established by the source antenna

arises from the uncertainty in the net power delivered to it. In turn, this uncertainty

reflects our lack of knowledge of the amplitudes and phases of the various reflection and

transmission coefficients in the power delivery system, as well as the uncertainty in

measurements of the power incident upon, and reflected from, the source antenna. Thus, we wish to compute more accurately the net power delivered to a standard transmitting antenna

from measurements of incident and reflected powers obtained with a dual directional coupler.

Our power delivery and measurement system can be represented by a four-port black box as shown

in Fig. 5.2. The port terminations and numbering are:

(1) power meter to monitor the forward (throughput) power,

(2) power meter to monitor the power reflected from the

standard transmitting antenna,

(3) source of the CW RF power,

(^) transmitting antenna.

The net power delivered to the transmitting antenna representing the difference between

incident and reflected powers is given by [1]:

P = P - P net inc ref

P

1 3M I ,r. „m2

2 (1 - |rj )

(1 - Ir2l') Is^i,!'

p

2 1 K,„ „m2

|g(S, Dp

|h(S, r)|^ . (5.1)

The symbols P. and P are, respectively, power meter readings at ports 1 and 2, r, and frepresent the corresponding reflection coefficients observed looking into power meters 1 and

2, S. . is the scattering parameter defined as the ratio of the complex wave amplitude emerging

from port i to that incident upon port j, g(S,r) and h(S,r) are functions of the system Sparameters and the reflection coefficients of ports 1, 2 and k [1]. In an ideal coupler, i.e, zero reflection coefficient for all coupler input ports and infinite directivity (S. = S =

S_ = S^^^ = S,^ = S^, = Sp_ = S-p = 0), and for a matched power meter at port 1 (r, =0), it can be shown [1] that g(S,r) = h(S,r) = 1. Unless the magnitudes and phases of the system S65
parameters and the reflection coefficients are well determined, g(S,r) and h(S,r) are not

calculable. The extent of deviation from unity is, therefore, taken to be an error contribution to the determination of the net power delivered to the standard antenna.

Although the degree of deviation from unity is a function of the system S and r parameters, it

is found to be, in general, less than ^%. [1].

To compute the net power given in (5.1), the terms S^^/S, _ and 1 /S„^

determined. Although the magnitudes of S , S^, and S^

need to be

could be measured with a network

analyzer, the system implemented at NBS for establishing standard EM fields is a selfcalibrating system which utilizes a standard flat-plate short and a matched termination. When

a short (V^ = 1 ) is placed at port 4, the ratio of power measurements ?„ and P gives

1 - ir,r

(1 M^' (5.2)

where A.(S,r) is a complex quantity much less than unity [1]. The second step in evaluating

the net power delivered to a transmitting antenna is to replace the short at port M with a well-matched power meter. The ratio of the two power measurements P and P^^ is

13

3^

1 - |rj

(1 ^ \U') (5.3)

where l\ is another complex quantity much less than unity [1]. From (5.2) and (5.3), a value

for |l/S„^| is obtained. In summary, we perform two powei—ratio measurements with a standard

short and a matched termination in order to determine Is^^^/S 3^+13 and

2H

The terms, A and Ap, involve the products of the system S parameters and reflection coefficients r. Since

the magnitudes and phases of A- and A- cannot be easily determined, the extent of deviation

from zero is, therefore, taken to be another error contribution. Moreover, the uncertainty in

the power-ratio measurements, Pp/P-i ^"^^ ^I'^^ii' ^'^'^ ^^^ uncertainty in the reflection

measurements r., f-, and r^, also contribute an error in the determination of |S-j^/S.-| and

The detailed discussion on this topic may be found elsewhere [1]. Thus, the net power

supplied to a transmitting antenna is determined from the absolute power measurements. P., Pp,

'2H ipp

and P

5.2 Near-Zone Gain of Open-Ended Rectangular Waveguide

The EM field measurements in an anechoic chamber are usually performed in the near-field

region of the transmitting standard antenna, and the approach used to establish the standard

field strength is to calculate the radiated field strength in the near-field region of the

transmitting antenna. The antenna considered for this purpose consists of a series of openended waveguides below 500 MHz and a series of rectangular pyramidal horns above 500 MHz.

The near-zone gain of an open-ended, unflanged rectangular waveguide is calculated from a forward near-field power pattern, which is determined from the conventional far-field power

pattern by use of the plane-wave scattering theorem [2].

The geometry of the open-ended rectangular waveguide is shown in Fig. 5.3. The E-plane

pattern, E (e), is predicted quite accurately by inserting the electric and magnetic fields of

the propagating TE mode into the Stratton-Chu formula and integrating over the aperture of

the waveguide [3]. Thus, we have

66
R ra^ - fl {1 + (g/k) cose + r[i - (e/k) cose]}

E^^^ ^E [ 1 + B/k + r(l - 6/k)]

sin(kb sine/2)

kb sine/2 (5.M)

2 1 /2 where the normalized propagating constant g/k for the TE mode equals [1 - (ir/ka) ] , with

k being the free-space wave number,, and r is the reflection coefficient of the TE mode from

the end of the waveguide. The constant A , which is related to the amplitude of the incident

TE.^ mode, will be defined later.

In the case of the H-plane fields, the aperture integration of the Stratton-Chu formulas

with the electric and magnetic fields of the TE mode neglects the fringe currents and,

therefore, produces much too broad an H-plane pattern. Using an accurate estimate of the

fringe currents on the x = ± a/2 sides of the rectangular waveguide from a numerical solution

to the electric-field integral equation applied to the open-ended rectangular waveguide [4], we obtain for the H-plane pattern

„ ,„, . r(cose + B/k) + r(cose - g/k) ^ „ -1 ,, . „,„v

E (e) = A [ ^ zf- + C J cosCka sine/2)

(Tr/2)^ - (ka sine/2) ° (5.5)

The constant A^, is related to A in (5.^) by

Ag = A^{(2/Tr)^[l + B/k + r(1 - g/k)] + C^} (5.6)

The constant C is calculated by equating the radiated power determined from the far-field to

the total input power determined from the TE ^ mode field.

Once the far-field power pattern of an open-ended rectangular waveguide is determined,

the plane-wave scattering theorem enables us to predict its near-field power pattern [5]. The near-zone gain of this antenna is then determined by integrals of its near-field power

pattern. The evaluation of the uncertainty of the near-zone gain will be performed by

comparing it with the experimental results [6],

5.3 Near-Zone Gain Calculations of Rectangular Pyramidal Horns

The approach used at NBS to establish a standard field at frequencies above 500 MHz

involves the use of a series of rectangular pyramidal horns. In deriving the near-zone gain

of a pyramidal horn by the Kirchhoff method, Schelkunoff accounted for the effect of the horn

flare by introducing a quadratic phase error in the dominant mode field along the aperture

coordinates [7]. Geometrical optics and single diffraction by the aperture edges yields

essentially the Kirchhoff results. The proximity effect in the Fresnel zone can also be

approximated by a quadratic phase error in the aperture field.

To improve Schelkunoff ' s equation by taking into account the reflection of the diffracted

fields from the horn interior and double diffraction at the aperture, the concepts of the

geometrical theory of diffraction are used to determine the on-axis near-zone gain of an Eplane pyramidal horn. Taking into account the preceding considerations, we obtain the

improved near-zone gain of a pyramidal horn [8]:

G = ^^R„ R„ (5.7)

,2 EH ttA where R and R are the gain reduction factors due to E-plane and H-plane flares,

L H respectively. The pertinent horn dimensions used in (5.7) are shown in Fig. 5.4. The factor

2 32 ab/(TrA ) is the gain of an in-phase field distribution uniform across one dimension of a rectangular aperture and cosinusoidal across the other.

67
The H-plane flare of the horn is given by [8]:

Tr^{[C(u) - C(v)]^ + [S(u) - S(v)]^} .^ „- " 4(u - v)^

where C and S are the Fresnel integrals defined as

C(w) - jS(w) = /q exp(-JTit^/2) dt , (5.9)

and their arguments u and v are defined as u=A+B, v=-A+B,

with

1 1 /2

(2A£')^/2 ^ "

i^ = rJLj^/(r + l^)

and r being the distance between the center of the horn aperture to the field point.

The E-plane factor R is given by [8]:

b

1 + COS<|) , Rg = 2"^ |exp(-jkilg cosct.^) + 2v(il'. ^ - ^^) ^ ~ ^2^ ' 4w

where

(5.10)

(5.11)

w = T-pr , and X,' = ri /{r + i ) . (2X ili)'^'^cos((t> /2)

ti h t, E o

The factor v(il, a) is given by:

vil, a) ^ - ""P^^^ ^°^"^

[1 - (1 - j) D] . (5.12)

with

D = C[(4k!l/Tr)^^^ cos(a/2)] - j S[ (iiki/ir) ^ ^^ cos(a/2)] .

The factor S is defined as

m

^2 ^iSl^'^'^E' I

' ^*o^ ^^^^i' ^ ' *o' I' ^*o^ ' ^^-^^^

where

f(d, e, e^) = v(d, e - 6^) + v(d, e + e^) , (5.14)

d = 28, sin(i(t) ) is the ray-path length between single and double diffractions, and m is the

largest integer less than it/2(j) .

The near-zone gain, (5.7), of a pyramidal horn is then used to calculate the radiated

field strength in the near zone of the antenna. The typical gain reduction factors R„ and R„

h n expressed in decibels are shown in Fig. 5.5. The evaluation of the uncertainty of the gain

reduction factors will be performed by comparing the theoretical values with the experimental

results [6].

68
5.4 Reflections from Anechoic Chamber Walls

The failure of the chamber to provide a true, free-space test environment affects the

measurement accuracies. The performance of a rectangular RF anechoic chamber can be checked

by measuring the relative insertion loss versus separation distance between a source antenna

and a receiving antenna [9].

Relative insertion loss is the ratio of power received by a receiving antenna or probe

for the initial test position to that received for different test positions. It is assumed

that the input impedances of the source antenna and probe and the power transmitted by the source antenna all remain unchanged. If the anechoic chamber were a perfect free-space

simulator, the relative insertion loss would vary with distance according to the free-space

transmission loss formula given by [7]

P^/P^ = ggg (X/i4Trd)^ (5.15)

where P is the net power transmitted by the source antenna, P the power received by the

probe, g the near-zone gain of the source antenna, g the near-zone gain of the receiving

s p probe, d the antenna separation distance in meters, and X the wavelength in meters.

Measured data are compared to the free-space transmission loss calculated using the

appropriate near-zone antenna gains. Disagreement between the measured insertion loss and

calculated transmission loss is a measure of reflections from chamber surfaces, assuming the near-zone gain calculations are exact for the separation distance considered. The measured

relative insertion loss versus antenna separation distance provides voltage-standing-waveratio (VSWR) data by means of a longitudinal probe scan. Rear-wall reflections and source-toprobe interactions are often resolvable at all frequencies, but reflections from the ceiling,

side walls and floor are difficult to identify at frequencies below 500 MHz because the VSWR

period is too long. Figure 5.5 shows an example of measured relative insertion loss with

calculated free-space transmission loss along the axis of the horn antenna.

REFERENCES

[1] Kanda, M. ; Orr, R. D. A radio-frequency power delivery system: procedures for error

analysis and self-calibration. Nat. Bur. Stand. (U.S.) Tech. Note 1083; 1985 August.

[2] Kerns, D. M. Plane-Wave Scattering-Matrix Theory of Antennas and Antenna-Antenna

Interactions, NBS Monograph 162; 1981 June.

[3] Stratton, J. A. Electromagnetic Theory. New York, NY: McGraw-Hill; 19^1.

C^] Yaghjian, A. D. Approximate formulas for the far fields and gain of open-ended

rectangular waveguide. Nat. Bur. Stand. (U.S.) Int. Rept. 83-1689; 1983 May.

[5] Yaghjian, A. D. Efficient computation of antenna coupling and fields within the nearfield region. IEEE Trans. Antennas and Propag. AP-30(1): 133-138; 1982 January.

[6] Kanda, M; Orr, R. D. Neai—field gain of a horn and an open-ended waveguide: comparison

between theory and experiment. Fifth International Conference on Electromagnetic

Compatibility, Institution of Electronic and Radio Engineers; September 29 to October

2, 1986; University of York, England,

[7] Schelkunoff, S. A. Electromagnetic Waves. New York, NY: Nostrand Reinhold; 1943.

[8] Jull, E. V. Errors in the predicted gain of pyramidal horns. IEEE Trans. Antennas and

Propag. AP-21(1): 25-31; 1973 January.

[9] FitzGerrell, R. G. Using free-space transmission loss for evaluating anechoic chamber

performance. IEEE Trans. Electromag. Compat . EMC-24(3): 356-358; 1982 August.

69
— ^^

5 m

Figure 5.1 Side view of the NBS anechoic chamber,

I I COUPLER

THRU LOSS

CABLE LOSS

,^^rzzr2 ^^^

Figure 5.2 Measuring the net power to a standard

antenna.

70
Figure 5.3 Geometry of open-ended rectangular waveguide,

71
uj

DC

06

I

GC

-2

Figure 5.4 Pyramidal horn dimensions,

-T

1 —1

1

' 1 1 —r-

"

Re - z:

-—

^

_i 1

R»

1 1 1 1 L_

— 12 3 4 5 1 Range, meters

Figure 5.5 Near-zone gain reduction factors, IL. and R^, of a typical pyramidal horn at 1,000 MHz.

72
m

COCO

3

c

o

CO

c

>

Theoretical

Measured

2 3

Separation, m

Figure 5.6 Relative insertion loss between a horn antenna (main

beam) and a probe, with the free-space transmission

loss curve fitted at a separation distance of 1 m. Frequency = 500 MHz.

73
Chapter 6. OPEN-FIELD MEASUREMENTS

The National Bureau of Standards offers a calibration service for field strength meters

and EMI antennas in the frequency range of 10 kHz to 1,000 MHz Cl]-[3]. The main part of a calibration consists of determining the "antenna factor" (K) which permits a receiver (RF

voltmeter) to be used with the calibrated antenna to make measurements of field strength. The

factor (K) can be used to convert the receiver dial indication in yV or dBpV to field strength

in liV/m or dByV/m. The types of antennas involved are basically loops for measuring the

magnetic field from 10 kHz to 50 MHz, dipoles for measuring the electric field from 25 to

1 .000 MHz, and monopoles for vertically polarized electric fields from 30 kHz to 300 MHz.

There are two independent techniques by which field strength can be evaluated. These are called the "standard field" method and the "standard antenna" method. The former consists of

generating a calculable standard field component which is determined in terms of the type and

dimensions of a transmitting antenna, its current distribution or net delivered power, the

distance from the transmitting antenna to the field point, and the effect of ground

reflections (if present). The latter method consists of generating an unknown field, but

measuring it with a calculable receiving antenna. The voltage or current induced in a standard antenna by the component of field being evaluated is measured. The field-strength

value is then calculated in terms of this induced voltage, the dimensions and form of the

receiving antenna, and its orientation with respect to the field vector.

At frequencies below about 50 MHz for loop antennas, a quasi-static near-zone magnetic

field is produced by a balanced single-turn transmitting loop of 1 0-cm radius. Above 25 MHz

for dipole-type antennas, a radiated far-zone electric field is produced and evaluated in

terms of the open-circuit voltage induced in a self-resonant receiving dipole. Between 30 kHz

and 300 MHz for vertical monopoles and small probes, an elliptically polarized electromagnetic

field is produced by a transmitting monopole above a 30 m x 60 m conducting ground screen.

The instrumentation used by NBS and the uncertainties for these calibrations are discussed in

this chapter. All the techniques described here for field-strength standards are applicable

only for steady-state RF fields with sinusoidal time variation. They are not intended for use with pulsed fields or other broadband applications.

6.1 Standards for Field Strength Meters, 10 kHz to 10 GHz

For most field strength meters, the first part of a calibration is checking the receiver

as a tunable RF voltmeter [4], [5]. This generally includes measurement of linearity (dial

indication vs. input level) and checking the internal step attenuators at several frequencies

[3]. A summary of these tests is given as follows:

Frequency Amplitude Calibration

Type of NBS Measurement range range uncertainty

1. Receiver indication 10 kHz-iJOO MHz 1 yV-IO mV ± 1/2 dB

as a function of ^100 MHz-1,000 MHz 10 y V-1 mV ± 3/4 dB

signal frequency, at 1 GHz-10 GHz 100 yV-lOO mV ± 1 dB

a few voltage levels.

2. Attenuation of the 10 kHz-1 GHz (0 - 80 dB) ± 1 /4 dB

step attenuators at 1 GHz-10 GHz (0 - 80 dB) ± 1 /2 dB

a few frequencies.

3. Receiver linearity, 10 kHz-1 GHz 10 yV-100 mV ±1/4 dB

which is independent 1 GHz-10 GHz 10 yV-100 mV ±1/2 dB

of frequency.

6.2 Magnetic Field Strength Standards for Loops, 10 kHz to 50 MHz

The standard field method is used here. The response of an electrically small receiving

loop antenna is proportional to the average normal component of magnetic field strength

incident on the antenna. At NBS, a calculable quasi-static magnetic field is produced for

calibrating these antennas, using a circular single-turn balanced transmitting loop. The

current in this loop of 1 0-cm radius is approximately constant in amplitude and phase around

the loop.

The receiving loop antenna being calibrated is positioned on the same axis as the

transmitting loop at a distance of 1.5 to 3 m. Figure 6.1 shows the NBS calibration setup.

74
The normal component of the magnetic field, averaged over the area of the receiving loop, is

given by [6] :

„._i;i [, . 15( !iJi,2] n ..^R^,"2 (6.1)

where H = RMS value of the magnetic field, A/m,

I = RMS current in the transmitting loop, A, r = radius of the transmitting loop, m, r_ = radius of the receiving loop, m,

^^2 2 2,1/2

Rq = (d + r^ + r^)

d = axial distance between the two loops, m,

k = 2it/X, and

A = free-space wavelength, m.

The current in the transmitting loop is measured with a vacuum thermocouple calibrated

with direct current. It is at the top of the loop winding, as shown in Fig, 6.1, and its dc

output is measured with a millivoltmeter . The RF/dc substitution error of the thermocouple is

< ]% at frequencies up to 50 MHz. Equation (6.1) is accurate within ± 0.2? if kR ^ 1 and

2 2 2 1/2

r r /R < 1/16. At higher frequencies, the correction term (1 + k R ) becomes appreciable

for the usual spacing of 1.5 to 3 tn used at NBS. The uncertainty of calibrating loop antenna

factors at NBS is ± 1/4 dB for frequencies up to 5 MHz, ± 1 /2 dB between 5 and 30 MHz, and ± 1 dB between 30 and 50 MHz.

While coaxial loops are normally used for calibrating purposes, the two loops can also be

positioned in the same plane. Co-planar loops are advantageous under certain conditions,

e.g., with some ferrite core antennas in which the core length is large. In this case, the

calibrating value of the magnetic field is half of that given by (6.1).

The calibration and subsequent measurement of magnetic field (H) are often expressed in

terms of the electric field (E) that would exist if the measurement were made in free space,

in which case E/H = 1 20it fi. When such a field-strength meter is used to make measurements

near the ground, the indicated value of the electric field is not necessarily valid. The same

is true for measurements made in the near zone of a transmitting antenna. However, the value

of the magnetic component (H) can still be measured correctly.

For calibrating loops or H-field probes at a higher field level, it is possible to use

the calculable magnetic field generated in a TEM cell (see chapter 3), or at the center of a flat multi-turn coil, or at the midpoint of a Helraholtz coil pair.

6.3 Electric Field Strength Standards for Dipoles, 25 to 1,000 MHz [1],[7]

The standard antenna method is used for this purpose. The magnitude of the electric

field component at a given point in a locally generated field is determined at NBS from the

open-circuit voltage (V ) induced in a standard half-wave receiving dipole. The field-site

instrumentation is shown in Fig. 6.2. The induced voltage is measured across the center gap

of the dipole, which is oriented horizontally and parallel to the electric-field vector of the

incident field. In using the standard antenna method, a plane-wave field is generated by a suitable transmitting antenna such as a log-periodic or half-wave dipole. The magnitude of

this incident field is measured with the standard dipole by the relation

E. = V /L _„ , (6.2) mc oc eff

where

E. = strength of the locally generated field, V/m,

V = open-circuit voltage induced in the standard

dipole, V, and

L „„ = effective length of the standard dipole, m .

75
The RF voltage picked up by the half-wave standard dipole is measured in terms of the

rectified dc voltage, as detected by a high- impedance Schottky diode connected in shunt across

the center gap of the antenna. The diode output is filtered by a balanced RC network, and

this dc voltage is measured with a high- impedance dc voltmeter. As shown in Fig. 6.3, the

RF/dc substitution of the detector network in the standard dipole is determined by applying a known RF voltage at 50 MHz across the standard dipole gap. The antenna rods making up the

dipole are removed for this measurement.

The high impedance of the diode "voltmeter" eliminates the necessity for a separate

measurement of the dipole impedance, since the source impedance of a resonant dipole is low

compared with that of the voltmeter. Another advantage of this approach is that the

measurement of E. is not affected appreciably by the presence of ground or other perturbing

objects. However, a subsequent measurement of field strength by the calibrated customer

dipole will be affected by ground due to impedance change of the customer's dipole, which is

loaded with a 50-f2 receiver. Also, the standard dipole used to measure E. lacks frequency

inc

selectivity, so it is not possible to perform the NBS calibration in the presence of strong

interfering signals.

The range of filtered dc output voltage used at NBS is normally 0.5 to 1.5 volts, which

is generally sufficient to avoid calibration uncertainty caused by ambient fields and

temperature changes of the diode detector. The response of the diode voltmeter is independent

of frequency up to about 500 MHz, with slightly increased response above this frequency due to

the approaching series-resonant frequency of the diode mount.

Effective length of a receiving dipole is a measure of the E-field intercept length,

analogous to the effective area of an aperture antenna. The effective length is derived in

terms of the current distribution on a transmitting dipole. The reciprocity theorem is

invoked in order to use the same value of L .„ for a receiving dipole. It should be noted

eff

that L „„ cannot be defined in terms of the current distribution on a receiving dipole. By

definition, the effective length of a transmitting dipole is

Moment of dipole current distribution

Leff Input current at the feed point (center) of the dipole

Assuming a cosinusoidal current distribution on an inf ini tesimally thin dipole, the

effective length of a half-wave dipole in free space is

^ff = ^/^ ^6-3)

However, the current distribution on a real dipole is not exactly cosinusoidal. An

approximate solution for the current distribution on a cylindrical transmitting dipole was derived by Schelkunoff [8], [9] and this solution is used for calculating L „. of the NBS

standard dipole. Also, in order to achieve self resonance of the dipole (zero reactance), it

is necessary to make its length slightly shorter than X/2. The required length for resonance

as derived by Schelkunoff depends on the dipole length-to-diameter ratio, and is

Required length L = (X/2) [1 - ^^°'^p^^_ ^ ] , (6.i4)

where D is the diameter of the standard dipole in meters.

The effective length of a thin dipole near resonance is given by

L „- = - tan(TrL/2A) (6.5) eff ir

If a half-wave dipole is shortened slightly to obtain zero reactance, the theoretical

resistance (R. ) depends on the length-to-diameter radio (L/D) as follows:

L/X at resonance Dipole ^ R.in

0.47 61 Q

0.48 66 n 0.49 70 Q

76

L/D ratio

50

500

50 ,000
The standard dipole sets used for field strength calibration at NBS are made of

cylindrical metal tubes which are 3 to 5 percent shorter than the free-space A/2. To the

first approximation, equation (6.3) can be used to calculate L „„ for these dipoles.

As indicated in Fig. 6.2, the antenna factor (K) of a customer's antenna or other antenna

under test (AUT) is determined by placing the antenna in the same field environment at the same position as the standard dipole, and using the following relation:

K = E. /V^^„

inc 50 f2

or (6.6)

^dB = 20 log E.^^ - 20 log V^Q^

where V^-„ is the voltage produced by the AUT across a 50fl load in volts, and E. is obtained

from the standard dipole measurement by means of (6.2).

Note that the possible impedance mismatch between the antenna and its load (50fi receiver)

is included in the value of K. Figure 6.4 is a graph of the theoretically expected antenna

factor for a thin A/2 dipole. Most often, the antenna factor may be expressed in decibels as follows:

dB/yV/m dByV

In this form, K is a dimensionless ratio given in terms of the incident E-field with respect

to 1 yV/m, and the antenna response across the load with respect to 1 yV.

With the K factor so determined, an unknown E-field can be obtained with the calibrated

antenna in terms of the measured pickup voltage at the 50-1^ receiver by

E = K V^Q^ , (6.8)

or

It could be noted that the customer's receiver, with attached antenna, can also be

calibrated as a system to measure the field strength directly.

A field-strength measurement made with the calibrated AUT may be in error if it is made

at a lower antenna height than that used during the calibration (about 3m), or if the

electrical ground constants at the measuring site are appreciably different from those at the

calibrating site [1]. This results from a change in input impedance of the customer's antenna

due to the proximity of the ground. A change of impedance does not affect the measurement of

V by the open-circuit standard dipole, but changes the mismatch between the customer's

antenna and a 50n receiver. However, if the customer's antenna is calibrated at a height

greater than 2A, the calibration results in essentially a free-space value. The error is

generally less than 10^ for heights greater than 0.5A. The error would also decrease if the

antenna load impedance (receiver input impedance) were high, and would approach zero as the

load impedance approaches «>. Also, if the receiver had a sufficiently large value of input

impedance, the calibration would be independent of the antenna polarization and of the ground

properties. Figure 6.5 is a graph of the impedance of a typical dipole, and Fig. 6.6 is a graph of the impedance of a typical loop for comparison.

The National Bureau of Standards usually determines antenna factor at an outdoor site (a

30-m X 60-m ground screen) which is provided with an aii—inflated all-weather cover. The

calibration uncertainty at the present time is about ± 1 dB.

6.4 Electric Field Strength Standards for Vertical Monopoles, 30 kHz to 300 MHz (Also

Standard-Field Method)

After considering several approaches for generating a standard (calculable) field to

calibrate vertically polarized antennas, the system chosen at NBS consists of a thin

cylindrical transmitting monopole over a 30-m x 60-m metallic ground plane. The field

77
strength is calculated in terms of the magnitude and distribution of the monopole current and

other factors, such as monopole height, horizontal distance from the transmitting monopole to

the field point, vertical height of this field point above the ground plane, and conductivity

of the ground system.

The basic arrangement of the NBS standard-field system is shown in Fig. 5.7. The height

of the transmitting monopole is adjustable, with a maximum height of about 3 tn . The

electrical height of this antenna is X/4 at 25 MHz, but only 0.0003A at 30 kHz. At

frequencies above 25 MHz, the antenna height is reduced to a X/H value. The base diameter of

the monopole used is about 1 cm. The monopole is excited through a coaxial cable from the

transmitting room located beneath the concrete ground slab.

The magnitudes of the three field components E , E , and H of a transmitting X/^

z p (p monopole above a perfect ground plane of infinite extent are given as follows, with the

geometry for these field shown in Fig. 6.8.

exp(-jkr ) exp(-jkr ) E = 30 I- [ — + — ] , (6.10)

z '^ r^ ""2

30 I„ , exp(-jkr,) . exp(-jkr^)

Lq H = ]j— (exp(-jkr^) + expC-jkr^)] , (6.12)

where

E = vertical E component, V/m.

E = horizontal E component, V/m,

H^ = magnetic (H) field encircling the monopole, A/m,

<p I-. = RMS base current of the monopole, A,

k = 2it/A, and

z, p, r, , and r„ = distances to the field point. m.

For frequencies near self resonance, the monopole base current is measured with an RF

ammeter consisting of a thermoconverter , which has been calibrated with known values of dc

current. At lower frequencies where the monopole input impedance is a high capacitive

reactance, the base current is calculated from Ohm's law in terms of the base voltage measured

with a vacuum-tube voltmeter and the theoretical input impedance. At very low frequencies, the

input impedance Z. = 1/(J(dC ), where the antenna capacitance may be calculated from

X 11 d. Schelkunoff 's equation [9]:

„ 55.63 h „ /c , '>^ ^a = iln(h/a) - 1

' P^ ^^-^3)

where h is the monopole height and a is the monopole radius, both in meters. For a 3~m

monopole 1 cm in diameter, C = 30.9 pF.

For a finite ground plane, the current on a vertical monopole will depart from the ideal

sinusoidal distribution of a filamentary monopole. This does not seem to affect seriously the

calculated values of far-zone field components. But the low-frequency, near-zone, quasistatic, electrical-field components so calculated will have greater uncertainty.

Examples of calculated standard-field data are shown in Fig. 6.9 for a 3-tn monopole

operating at 25 MHz, for which the electrical length is A/4. The base current used in this

case is 1 A. Vertical and horizontal components of the electric field are plotted versus horizontal distance from the antenna, for a height 1.22 m (4 ft.) above the ground plane.

Similar curves are shown in Fig. 6.10 for the same antenna at a frequency of 200 kHz, which

corresponds to the near-zone case.

78
If a transmitting monopole is electrically short, the current distribution may be

considered triangular approximately. The field equations under this condition are more complicated than those given in (6. 1 0)-(6. 1 2) , but can be programmed [10]. It should be noted

that the EM field values in the half space above ground are the same as those in each half

volume of a center-fed A/2 dipole in free space. The input impedance of a monopole above

perfect ground is half that of a dipole in free space. The power required to generate a given

field strength is half that required for a dipole, but the radiated power goes into half the

volume, so the field is the same. Measurements of Z. at the NBS ground-screen facility with in ^ J a commercial impedance meter were performed to check the theoretical values from 0.5 to 50

MHz. Measurements of the monopole capacitance were made at lower frequencies with a commercial Q meter. Also, checks have been made of the calculated electric field versus measurements with a small active calibrated field-strength probe. The agreement found between

the various techniques was within 1 dB at all frequencies checked.

A metal ground plane having a thickness of several skin depths, so that the ground

currents will not penetrate through, exhibits essentially infinite conductivity for the

frequency range covered here. The dimensions of the NBS ground screen (30 m x 50 m) are several times the height of the monopole used (up to 3 m) in order to minimize the effect of wave reflections from the edges of the plane. At low frequencies where the ground screen is

electrically small, it is difficult to achieve capacitive coupling to the reinforced concrete

slab or to the surrounding earth, even though the underground transmitting room has a large

ground rod. Therefore, further experimental testing is required of field uniformity versus azimuth angle, and field-strength reduction versus distance from the transmitting antenna.

For the purpose of calibration, the antenna factor (K) of the AUT is determined by

immersing it in a calculated vertically polarized field (E ) about 20 m from the NBS

transmitting monopole. A coax cable is usually attached to, and considered part of, the

antenna when calibrating K. The equation used is the same as that for calibrating dipoles,

namely (5.6) or (5.7). The theoretically expected antenna factor of an electrically-short

monopole above a perfect ground is given by

Z K = (1/h^^^)|[l - ^

]\ , (6. in)

where h is the effective height of the receiving monopole in meters, and Z is the monopole

input impedance in ohms.

For example, for a 1 -m whip of 0.5-cm diameter at frequencies below 10 MHz, h = 0.5 m and C = 11 pF. In this case Z = -j l4470/f,MHz and K = 579/f,MHz, where f.MHz is the

a a operating frequency in MHz.

It can be seen that the antenna factor of a short monopole (or dipole) is inversely

proportional to frequency, while the antenna factor of a resonant monopole (or dipole) is

directly proportional to frequency.

An unknown electric field strength (E) can be determined with a calibrated monopole by

measuring its output voltage with a 50-fi receiver or spectrum analyzer, and using (5.9).

The calibration of monopole antenna factors at NBS is performed at the outdoor groundscreen site for frequencies from 30 kHz to 300 MHz with an uncertainty of ± 1 dB. For

frequencies up to 10 MHz and for monopoles or probes having a height less than 0.5 m, these

calibrations can also be performed in a large TEM cell with the same uncertainty.

REFERENCES

[1] Greene, F. M. NBS field-strength standards and measurements (30 Hz to 1000 MHz).

Proc^ IEEE, vol. 55(6): 970-981; 1967 June.

[2] Taggart, H. E. Field strength and RFI standards at the National Bureau of Standards.

IEEE Int. Symp. on EMC. 1968 July; Seattle, WA.

[3] Taggart, H. E. ; Workman, J. L, Calibration principles and procedures for field

strength meters (30 Hz to 1 GHz). Nat. Bur. Stand. (U.S.) Tech. Note 370; 1959 March.

79
1^1

[5]

[6]

[7]

[8]

[9]

[10]

The radio frequency interference meter. NAVSHIPS 98180, U. S. Navy; 1962 July.

Clark, D. R. An investigation into existing calibration methods of field intensity

meters. RADC-TR-64-527 ; 1965 January.

Greene, F. M. The near-zone magnetic field of a small circular-loop antenna,

Nat. Bur. Stand. (U.S.). 71C(4): 1967 October-December.

Res

IEEE Standard, Methods for measuring electromagnetic field strength for frequencies

below 1000 MHz in radio wave propagation. IEEE Std. 302; 1969 August.

Greene, F. M. ; Solow, M. Development of very-high-frequency field intensity standards.

J. Res. Nat. Bur. Stand. (U.S.). 44: 527-5M7; 1950 May.

Schelkunoff, S. A.; Friis, H. T. Antennas, Theory and Practice

Wiley and Sons; 1952.

New York, NY: John

Jordan, E. C. Electromagnetic Waves and Radiating Systems. Englewood Cliffs, NJ

Prentice-Hall; 1950.

DC Milli- Voltmeter

RF Signal

Generator

^Vacuum thermocouple

Heater element

Transmitting loop

Balun

J

Receiving loop

Figure 6.1. Diagram of the NBS measurement system for calibrating

loop antennas.

80
DC<

III -^ > _) nr SBS

LU LU -J

DC Q. < 1 C^

CO z o

in

O

DC

LU Ul

I—

LU

Oz< b^ Q ox LU > Q. o ^ Q

LOW

PASS

FILTER oc

LU o>

Qi

O

=: UJ

LU ^.1 OX

DC2

70o o

CO ^^ u_ o

DC CNJ

a;o

cl

<T3 4-) coo

JCenc

4->

(Jcoo

i-J c

(/) ISI c nr

q; c •M o •r- c t/i .—I D O

r— 4-) • r- LO

U- CM

0)u3

CO

81
CL

LLI 1-

111 -^ 2 CL

cc hN

III o >-

<:

m -J CL < u CO z

QDC <

ooo

in I k

CD

CO nr

W-PA

ILTE O^

_j; k

UJO

M

ccx

8^

CO o

in

li. OC

CO

LU

I—

LU C^ o

V.\fV/«#' o o

in in

3.J- -L =L 3 Li. 2co

LU> O

CLD

CL<QZ OoQ

LU

CL

CO<

LU cc

0)>oc

enc

COEco

i-> "3

*-> C

OJE • ^ OJ

(_ ,

4->

O

(/I Cl c — —' T3

on3

OC

82
40

GQD

o

UJ

. 30 - \Leff/\ 50 ; va/v 50

20

1—I

m

= (0.025) (F in MHz)

K in dB=20 log(K)

= 20 log (F in IV1Hz)-32.0

100 200

FREQUENCY, MHz

1000

Figure 6.4. Theoretically expected antenna factor of a thin X/2 dipole and 50 n receiver, excluding cable loss and balun loss.

83
2000

1000

COE

o

-1000

-2000 -

1000 2000

R, ohms

Figure 6.5, Free space impedance of a dipole antenna, length/diameter

ratio = 2000.

84
2000

J— j__

Max X=15,000Q

1000

CO

E

o

-1000

-2000

R at first resonance

=29,000Q

'*-T

=°-''

Max (-X)=-14,000Q

I I 1000 2000

R, ohms

3000

Figure 6.6. Free space impedance of a loop antenna, circumference/wire

diameter = 200.

85
Transmitting

Monopole

Monopole

to be

Caiibrated

<r)

^) Thermocouple

IT

DC

Voltmeter

it 1

RF

Source

KD

R.F. Voltmeter

Ground Screen

Concrete Slab

i^ or J en on_ ___^_J ^

Calibrated

50 Q Receiver

Spectrum

malyzer

Added Capacitance at Low Frequencies

Underground Transmitting Room

Figure 6.7. Field site instrumentation for calibrating monopole antenna factors.

86
To a>^^

Field Point

-///'/'/V/'/V/'/i /'/V/V/z/V'/'//- v^;^/^^/.

Image

Antenna

Ground Plane

>-To oo

Figure 6. 8. Geometry for the EM field of a thin X/4 transmitting monopole

antenna
0.1

Antenna Height: 3 m

Antenna Current: 1 amp

Frequency: 25 MHz

12 15 18 21

Horizontal Distance from Antenna, Meters

Figure 6.9. Theoretical electric field of a \/^ transmitting monopole
100

o

o

10

E

>

c

o

S.

o

(/)

"D

C

D

O

O

CD

>

O

<

^1.0

T3

0.1

1 i 1 1 i 1 —

^mt E^ (Horizontal Component) —

^" 1

\ Antenna Height: 3 m

*^^

\

\ Antenna Current: 0.015 amp

-\ \ Frequency: 200 kHz —

E.

_ __

—

V- ^x-^z^it d = 3 nn —

- \ \ E^ (Vertical Component)

:

— Ex\ \ —

1 1 \ 1 >\ 1 1 12 15 18 21

Horizontal Distance from Antenna, Meters

Figure 6.10. Theoretical electric field of a 0.002 X transmitting monopole

89
Chapter 7. EM PROBE DEVELOPMENTS

In order to measure quantitatively the EM environment, a small antenna or probe with an RF detector is often used. After sensing the RF field, the probe produces a dc voltage output

which can be processed through the associated instrumentation to indicate the field level.

Currently, there are two kinds of probes available, one sensing the electric-field strength

and the other the magnetic-field strength.

Two types of instrumentation are normally used in a receiving system: one is the

conventional field-strength meter consisting of a tunable meter or spectrum analyzer and a single antenna, which must be oriented for the desired field polarization. This kind of

receiver generally offers high sensitivity and selectivity, and is not designed as a portable

instrument for making quick surveys of the EM environment. The receiver thus acts as a frequency-selective voltmeter. The required conversion factor between the antenna pickup

voltage (volts) and the electric-field strength (volts per meter) is the antenna factor. It

must be determined experimentally for each antenna used in the system, at each frequency and

orientation by calibrating the antenna in a known standard field. The other type of

instrumentation employs a nontunable sensor characterized by lower sensitivity but broad-band

response. The antenna involved in this latter type preferably has a flat response over a wide

frequency range and a nondirective pattern. It is generally designed for measuring relatively

high-level fields exceeding 1 V/m, such as the region close to a transmitting antenna. The

frequency of interest is generally above 300 kHz. Thus, it can conveniently be used to assess

a microwave hazard environment.

7.1 Earlier Models

The program carried out at NBS in the past was to develop various probes in the frequency

range of 100 kHz to 18 GHz for measuring electric-field strengths of at least 0.1 V/m, or measuring magnetic-field strengths of 0.25 mA/m or more. The design principle of early probes was primarily based on temperature rise due to the absorption of RF energy [1]. Field

strength was indicated either by the changing color of a liquid-crystal material, the change

of resistance in a lossy dielectric, or by a glowing neon gas inside a miniature glass bulb

located at the center of the sensor. Other types of probes developed at NBS include centerloading a short dipole with a miniature incandescent bulb about 2-mm long or a thermocouple

heater [2]. The current induced in the bulb or heater by the field was then used as a measure

of the field strength. All of these early models lacked adequate sensitivity, quick reaction

time, good stability for calibration over a long period of time, dependable corrections for

variations in ambient temperature, and thus satisfactory measurement repeatability.

Another early probe developed at NBS was an "active" antenna system consisting of three

orthogonal dipoles, each sensing one component of the field (x, y, or z) [3]. This

arrangement avoids requiring physical rotation of the probe. Because the dipoles are perpendicular to each other, the mutual-coupling effect is relatively insignificant. This

system was designed for the frequency range of 15 kHz to 150 MHz. The dipoles, all

electrically short over the entire frequency range, are connected to a conventional tunable RF

receiver by a fiber-optic link. The antenna pickup is amplified first and then applied as modulation to the infrared (IR) output of a high-speed light-emitting diode (LED). The system

is considered active in the sense that the RF signal is amplified before the detection or modulation process. The modulated IR signals are guided through glass fibers to avalanche

photodiodes in the metering unit. These glass lines provide essentially perfect electrical

isolation between the sensing antennas and metering unit. Photodetectors recover the total RF

modulation from the IR carrier for input to the RF receiver. The readout indication at each

frequency is proportional to a single component of the probed electric field. A root-sumsquare (rss) circuit is also incorporated in the system to give the total magnitude of the

three field components.

The most important advantage of this probe is that complete signal information can be

recovered, including not only the signal strength but also frequency and phase

characteristics. The sensitivity of this system is also adequate. It may be noted that the

probe sensitivity decreases rapidly above the designed upper frequency limit of 150 MHz with

no unwanted enhancement at the dipole self-resonant frequency. One of the disadvantages is

the rather low dynamic range. In fact, both the tangential sensitivity and linear dynamic

range depend on the tuned frequency, noise figure, and bandwidth of the auxiliary receiver

included in the system. Another disadvantage is the limited upper frequency caused by the

limitation of LED speed. [However, heterodyne conversion employing an RF local oscillator

inside the active dipole could be used to achieve higher effective signal frequency, but this

approach has not been pursued.] In addition, the switching speed between measurements of

90
three field components is relatively slow because only one RF receiver was used for all three

dipoles. The level of spurious responses and intermodulation distortion is also rather high.

7.2 Recent Models

One of the recent probes developed at NBS is the broad-band, isotropic, real-time,

electric-field sensor (BIRES) [4], [5]. This probe covers a much broader frequency band,

typically from 10 MHz to 1 GHz, also using three orthogonal dipoles. Each dipole is

fabricated by depositing a thin film of metal alloy with varying resistivity on a glass rod.

The alloy, which consists of approximately 70 percent nickel, 15 percent chromium, 10 percent

iron, 2 percent titanium, and others, has a high resistivity and low temperature coefficient.

The glass rod, 15-cm long and 0.7 cm in diameter, serves as a substrate for the deposited

film. The antenna becomes a half-wave dipole at approximately 1 GHz. The required resistive

loading is about 5 kn/m at the center of each dipole, 10 kfl/m at midpoint between the center

and ends, and infinite at the ends of the dipole. It is possible to calculate the required

thin-film thickness. Typically, this is about 2M0 nm at the center, 100 nm at the midpoint,

and zero at the ends.

Because the tapered resistive loading makes the internal impedance per unit length a proper function of position along the dipole, the resulting current distribution on the

antenna is a pure outward-traveling wave. Hence the probe has linear amplitude and phase

responses over a broad frequency band, and as such may be used to measure fast time-varying

signals with minimum pulse-shape distortion. The probe also has a filtering action so that EM

signals outside the designed frequency range will be rejected, preventing out-of-band

responses. In addition to amplitude, phase, and frequency, the probe also provides

polarization information of the EM environment.

The BIRES probe uses metal coax cables to convey RF signals from the antenna to the

metering unit. Thus the electrical isolation of the antenna and resulting isotropy is not as good as the other probes discussed previously. Because the RF signal voltage delivered to the

conventional 50-n receiver from the dipoles is not independent of signal frequency, an RF

shaping amplifier is incorporated in the system for frequency compensation. The tangential

sensitivity and linear dynamic range also depend on the signal frequency and bandwidth of the

auxiliary receiver used.

Quite often, the regions being measured are close to radiating sources. In such cases,

the field structure is very complicated, including reactive (stored) and real (propagated)

components, standing waves, unknown phases, and unclear field polarization. The most

practical manner of surveying this kind of field environment is to use isotropic RF probes,

independent of orientation and direction of wave propagation, as hazard meters. It is

important that the probe be small in size and thus be able to resolve the spatial variations

in field strength. Futhermore, the field being surveyed should not be perturbed substantially

by the operator or equipment associated with the measurement. The end result of meeting these

requirements is represented by the NBS development of an "isolated" probe system [6]. A typical response at 20 dBV/m (10 V/m) of the NBS isotropic electrical-field monitor (EFM-5) is

shown in Fig. 7.1. A family of NBS designed electric-field strength meters using isolated

probes has been adopted as commercial meters by private industry.

The probe consists of three short dipoles mounted orthogonally in three notches cut near the end of a plastic tube. The dipoles are then embedded in a form sphere 10 cm in diameter.

The difference in transmission line between this probe and the BIRES discussed earlier is that

this probe unit employs three high-resistance plastic twinleads inside the dielectric handle

to bring the dc voltage to the receiver. These lines also act as RC low-pass filters for the

demodulated signal coming from the dipole. Nonlinear circuitry included in the metering unit

can be used to give the rss value of the three perpendicular field components at the measuring

point, to represent the total effect of all electric field components, all polarizations, all

arrival directions, and all frequencies within the passband of the antenna/filter combination.

The unit also includes switches to permit measurement of a single field component only,

allowing a choice of measuring either the average or peak amplitude of modulated signals.

Typical applications for this probe system are 1) measurement ef possible RF hazards caused by

diathermy equipment, industrial RF heaters, plastic sealers, and near fields of transmitting

antennas; 2) survey of ambient fields for low-level RF pollution caused by AM, FM, TV, CB, and

other broadcasting services; 3) check of EM fields in sensitive areas containing electroexplosive devices or flammable fluids; and ^) check of field strength in sites having

instruments which may be degraded by the presence of RF radiation.

91
Although the isolated probe has a fairly flat response over the designed frequency range

of 0.2 MHz to 1 GHz, it is found that it also responds to higher frequencies. Therefore, RF fields due to harmonics and other frequencies all contribute to the meter indication. Also,

the measurement accuracy is normally reduced for pulsed fields that have pulse widths lower

than 0.3 ms, because of the limited charging time of the RC filter line included in the

system. In addition, a possible erroneous increase in meter indication occurs when measuring

multifrequency fields (see chapter 9).

Another isotropic probe (MFM-10) for measuring primarily magnetic-field intensities was developed, based on the same principle, and incorporates much of the instrumentation used for

the isolated electric-field probe described above [7]. Three small orthogonal loops were used

instead of short dipoles. It provides neai—zone measurement of magnetic fields over a dynamic

range of 0.1 to 12 A/m with a relatively flat response in the frequency range of 300 kHz to

100 MHz. It also is capable of measuring each of the three orthogonal field components in

addition to the total field magnitude.

The mechanical and electrical configuration of an individual loop in this probe are illustrated in Fig. 7.2. The internal portion of each loop is composed of 5 turns of wire

with a detector at the loop center and a filter capacitor C connected between the two ends.

The diode, D , provides rectification of the RF signal induced in the loop. The diode

shunting elements, R, and C. , help produce a flat frequency response. The values of R., C,,

and C are critical to the proper shaping of this response. A normalized response curve given

in Fig. 7.3 shows that the maximum variation with frequency is less than ± 1 dB.

Two subsystems have been developed to be used with any of the broad-band isotropic

probes. One of them uses the fast Fourier transfrom (FFT) technique [8]. During each

measurement, the total amplitude of antenna pickups over a given frequency band is recorded on

a high-speed tape recorder in the time domain. These recordings are later analyzed with a computer by FFT processing to obtain a three-dimensional display of field strength versus frequency, with time as a third parameter. This device, therefore, enables an analysis of

changes of the spectrum occupancy with time. Another subsystem provides a microprocessor

control for rapid data acquisition of three orthogonal field components and the total field

amplitude [9], [10]. Both of these approaches require previous calibration on the antenna

factors involved.

7.3 Future Models

A shortcoming of most isotropic broad-band probes is the relatively long response time of

the RF sensor. It would be impossible or difficult to measure directly the peak-to-average

ratio of a modulated field or the momentary maximum envelope intensity in the beam center of a scanning antenna. It is often advantageous or desirable to have the capability of observing

the modulation on a signal being measured. In addition, the biological importance of

measuring peak levels of RF pulses has not been established partly because the measuring

instrumentation is not yet commercially available. To make up for this shortcoming, a new system employing laser diodes, single-mode fiber-optic guides, and optical modulators to

replace the conventional coax cables in any of the isotropic broad-band probes has been

developed, fabricated, and tested under laboratory conditions. To date, capability of

measuring electric fields of about 0.1 V/m at 1 GHz with 50 kHz bandwidth [11] and about 1 mV/m sensitivity at 10 kHz with 3 kHz bandwidth [12] has been demonstrated. These systems

preserve the amplitude and phase information of the electric field, and reduce the measurement errors due to antenna lead pickup and field distortion. Present effort is being directed

toward improving system performance by reducing the noise inherent in the probe system in

order to expand the measurement bandwidth and by improving the stability against thermal

drift. Increased use of optical fiber and integrated optics technologies will improve system

performance and lead to lighter weight and more compact transmitter/receiver modules. In view

of promising potentials of this system, design considerations of some of the important

components included in the system are outlined in Section 7.5.

To date, all the practical field-strength meters measure either the electric field or magnetic field separately. A new probe is being developed at NBS to measure the electric and

magnetic fields simultaneously so that an EM environment may be characterized more completely

[13]. This type of probe is particularly needed for measuring the near field where the

magnitude and phase angle of the wave impedance are unknown, and the electric- and magneticfield vectors are not necessarily orthogonal to each other nor in the same time phase. Thus

the new device is intended to measure not only the polarization ellipse of the field vectors

92
in a near-field environment, but also the time-dependent Poynting vector to indicate the

energy flow. The probe consists of three electrically isolated doubly loaded loops with the

capability of measuring the sum and difference of the detected voltages at the two opposite

ends of each loop. A diagram of a single loop with equal loads of Z is shown in Fig. ?.'<. Currents developed in the loads correspond to the electric-dipole and magnetic-loop responses.

More precisely, across one load, the electric response adds to the magnetic response, while

across the other load, the electric response subtracts from the magnetic response. To

separate the currents, a 0°/l80° hybrid is used to obtain the sum and difference of the

currents. Thus, the sum current I gives a measure of the magnetic field, whereas the

difference current I. gives a measure of the electric field.

A

The calculated sum and difference currents in a loop with a load impedance of 200 Q are presented in Fig. 7.5. The real part of the currents increases with frequency up to about 200

MHz, The magnetic-loop current is larger than the electric-dipole current up to 100 MHz,

while the electric-dipole current becomes more dominant above 100 MHz.

An experimental loop model with a radius of 0.16 m and wire radius of 0.02 m is shown in

Fig. 7.6. This loop is doubly loaded with 200 fi by using ^1:1 baluns and 50-Q resistive loads.

Zero-bias Schottky diodes are used as detectors, with high-resistance plastic transmission

lines connecting the loop and high-impedance dc voltmeter. Measured values of the real parts

of I and I for this loop placed in a TEM cell with a known incident electric field of 1 V/m

are included in Fig. 7.5. Although there is some discrepancy between the theoretical and

experimental results, which may be associated with the balun impedance, the preliminary

results indicate the validity of the theory.

Figure 7.7 shows the real parts of the magnetic-loop and electric-dipole currents as a function of load impedance at 10 MHz. It reveals that there is a critical load impedance, for

example 260 Q at 10 MHz, for which the two currents are equal. Below this critical impedance,

the magnetic-loop current is greater than the electric-dipole current. Above this impedance,

the reverse is true. Critical load impedance has only a slight frequency dependence, ranging

from 200 to 260 a for the frequency range of 1 to 100 MHz [131.

7.4 Diode Model

Some of the probes described earlier use a diode to convert the electric field being

surveyed to a DC voltage. The main advantage of including a diode is to make the frequency

response of the probe very flat, so that the system can be used as a portable and compact

hazard meter. Otherwise, since the input impedance of a probe (short dipole) without a diode

is mostly capacitive, the antenna factor for such a probe is, in accordance with (6.1M),

frequency sensitive. It would be difficult to determine the unknown field strength without

also detecting the frequency by a spectrum analyzer.

A beam-lead Schottky-barr ier diode is usually chosen for this purpose. It has a good

high-frequency performance due to a small junction capacitance, high sensitivity, and low

noise characteristics. When an electrically short dipole is terminated with such a diode, the

effect of loading may be analyzed by the simple Thevenin's equivalent circuit shown in Fig.

7.8, where the induced open-circuit voltage V.(t) at the diode terminal is given by

h^'^ - ^inc(^) ^ff ' ^'-'^

with E. as the incident electric-field component parallel to the dipole, and L ._ the

effective length of the dipole. The element C in Fig. 7.8 is the equivalent driving-point

3

capacitance of the short dipole, and the parallel combination of capacitance C . and nonlinear

resistance R represents a simplified model of the diode.

For an electrically short dipole, the effective length and the driving-point capacitance

are given respectively by [1^4]:

^eff = Q - 2 - an i1 "" • ^^-^^

93
2tt e h ^"^ ^a =

^ - 2 . ^n i^ ^ • (7-3)

where h is the half physical length of a dipole in meters, e^ is the free-space permittivity

in farads per meter, Q is the antenna thickness factor defined by fi = 2 iln(2h/a), and a is

the dipole radius in meters. For example, when h = 0.02 m, a = 2.8^(10) m, we have fi = -2

U.50, L „„ = 1.94(10) m, and C = 0.10 pF. Note that the symbol Q should not be confused

ell a with the unit for a resistance.

The current flowing through the nonlinear resistance R of the diode may be characterized

by its v-i relationship

i(t) = I^[exp(a V^(t) - 1] (7.4)

where I is the saturation current, which is assumed in this case to be 2(10) A, V (t) is

the voltage in volts across the diode junction, a = q/nKT = 38 V , q is the electron charge

-1 9 [1.6(10) C], n is the diode ideality factor [~ 1.05], K is the Boltzmann's constant

-23

[1.38(10) -^ J/°K], and T is the absolute room temperature [- 290°K].

At frequencies approximately higher than 10 MHz, we obtain the detected dc voltage

averaged over complete cycle as [15]

2 2 V. ^ a L -- E V - - " r 1 ]2 _ _ eff inc , . o

'

4 h + c /c ^ ' 2 ^^'^'

° ' ^d^^a 4(1 + C,/C r

d a

for a small sinusoidal induced steady-state voltage V. [SI V]; and

V = —!^i— = - !:^£L_!iii£ (7.6)

1 + C^/C 1 + ZJZ da da

for a relatively strong V.[> 1 V].

The frequency response of V with E. = 1 V/m for a sample probe and diode system is

shown in Fig. 7.9, where the analytical solutions are obtained by solving a first-order

nonlinear differential equation in accordance with the equivalent circuit shown in Fig. 7.8,

while the numerical solutions are achieved by using an approximate time-stepping difference

equation [13]. From Fig. 7.9, we see that the frequency response is indeed very flat. The

inverse of these curves represents the antenna factor for the entire system.

7.5 Application of Electro-Optic Techniques for Measuring EM fields

Electro-optic techniques for EM field measurements offer several advantages over

conventional measurements. The main advantages are due to the use of non-electricallyconducting fiber-optic links between the field probe and the system readout. The dielectric

link produces minimum perturbation of the field being measured and provides immunity from EMI

in the data transfer. It also enables field measurements to be made in areas where high

voltages or hazardous environments would preclude the use of electrically conducting leads.

The large information-carrying capacity of the optical fibers allows the phase and amplitude

of the test fields to be recovered and broad bandwidth operation. However, these advantages

are not without some drawbacks. The principal one is that the production and detection of the

optical carrier signal introduces considerable noise in the system. A less severe restriction

is that the optical interaction length necessary to obtain adequate sensitivity in presently

available electro-optical materials limits the frequency response to about 10 GHz for low

field strengths of interest.

94
The basic electro-optic field measurement system is shown schematically in Fig. 7.10.

Light from a laser source is launched into the uplink optical fiber and propagates to the

modulator. Voltage from the antenna is applied to the modulator and changes its

transmittance, causing the light intensity in the downlink fiber to vary. At the receiver the

light is detected by a photodiode and the intensity fluctuations generate a time-varying

electrical signal that is processed by the signal analyzer. The feedback element is often

necessary to control the polarization state of the light at the transmitter end of the system

in order to compensate for stress or thermally induced birefringence in the fiber or modulator

respectively. Such compensation is necessary when the modulator output is sensitive to the

input polarization state of the light.

The design and use of an electro-optic probe involves the interaction of all the

principal components and must be approached from a systems perspective. Trade-offs between

sensitivity, dynamic range and bandwidth must be made. A lower limit on the physically

detectable field is set by the antenna factor, voltage sensitivity of the modulator, and

system noise. The power levels and wavelength of the optical carrier depend on the

susceptibility of the modulator crystal to photon damage, the signal processing requirements

(e.g., streak cameras), and available sources.

MODULATORS Probably the most important component of the optically-sensed probe is the

modulator. A large effort has gone into developing modulators for fiber-optic communication

links. Most of the resulting technology is applicable to the fabrication of field probes.

Table 7.1 lists the most promising modulators by type along with some developmental types and

their basic characteristics. In all cases, an electric field induced across the material or device causes a change in its optical properties. For the majority, the change is in the

relative refractive index along different crystal axes (Pockels effect). This induces

birefringence that alters the light propagation characteristics. For best sensitivity and

linearity, the modulator should be biased so that approximately 50/J of the light is

transmitted for zero applied field. Since one objective in using the fiber-optic link is to

eliminate the use of conductors to the probe, the appropriate bias point should be maintained

passively (inherently built into the device) or by controlling the input polarization state at the optical transmitter. The operation of a modulator is depicted in Fig. 7.11 which shows

the change in transmittance, d(I/l^), for an applied voltage dV. The modulator can be

conveniently characterized by its V , the voltage required to switch the light from full-off

IT to full-on, and the slope of the curve at the bias point. These values are also included in

Table 7.1 for several representative modulators reported in the literature.

One particularly attractive modulator is the Mach-Zehnder interferometer fabricated using

optical, integrated-circuit technology on LiNbO^ crystals [12], [20]. It has high sensitivity

and can be passively biased by making one interferometer leg one quarter optical wavelength

longer than the other. However, in its conventional configuration on z-cut crystals giving

the best sensitivity, it has recently been shown that the material large pyroelectric

coefficient makes it difficult to maintain the proper bias point with temperature variations

[21].

Modulators that provide complementary outputs have some advantages over single output

devices. By using separate downlink fibers for the two channels and subtracting the output of one channel from the other after detection, the EM signal of interest is doubled.

Furthermore, if the phases in the two downlinks are carefully matched, the noise in the

optical source can be subtracted out. Those modulators with the potential for complementary

outputs are identified in the table. Unfortunately, they all suffer from having lower

sensitivities or, as in the case of the directional coupler, are difficult to fabricate with

the appropriate passive bias point.

The sensitivity or V voltage of a modulator is, in general, a product of the electric

IT field strength induced in the crystal and its interaction length with the optical carrier.

The reason that optical, integrated-circuit devices show very good sensitivity is that because

of their small size and electrode spacings of only a few micrometers, high field strengths are obtained. VJith the high field strengths, short interaction lengths of a few millimeters and

high-frequency response are also obtained. It should be noted that although traveling-wave

structures have been fabricated which can modulate carriers at frequencies above 10 GHz [22],

such devices may not be appropriate for field probes since the traveling-wave electrodes would

disturb the dipole characteristics.

95
RECEIVERS A second important component in the electro-optic probe is the receiver at

the downlink end of the system. It consists primarily of a photodiode and amplifier. If

complementary outputs are used, it also contains the channel subtraction network. For many

applications, the high signal frequencies and the shallow modulation depth of the optical

carrier demand the highest performance available for the receiver elements. The primary

criterion for adequate performance is that the receiver maintain as high a signal-to-noise

ratio as possible.

Several sources of noise in an electro-optic system need to be considered for field

probes. These are listed, together with some suggestions for minimizing them, in Table 7.2.

The reduction of the noise in the receiver requires that the detector and amplifier be

carefully matched, preferably designed to operate together and fabricated into one monolithic

structure. Detailed discussions of receiver design and noise characteristics can be found in

the literature [23]. For our purposes, a brief comparison of an avalanche photodiode (APD)

and a PIN photodiode is instructive. The analysis is outlined in Table 7.3. In both cases,

the total gain is 50, which is typical of available APDs. The power in the optical carrier is

chosen to be 50 microwatts, which is low enough to limit optical damage in a typical LiNbO^

optical integrated-circuit device. The modulation depth. A, is typical for a field strength

of 1 V/m using a dipole antenna with 15-cm-long elements. As indicated, the thermal noise in

the 50-fi input resistance dominates in the PIN receiver, while shot noise dominates in the

APD. This is because in the PIN the thermal noise is amplified by the gain, while in the APD

the gain occurs prior to the load resistor. The signal-to-noise ratio for the APD is thus

superior to that for the PIN receiver.

When shot noise dominates, the signal-to-noise ratio can be approximated by

2 2 -^^^^c

I / I = (7 7)

^s ^ n 2eB F(m) ' ^'''' n

where the various quantities are defined in Table 7.3. Three parameters can be changed to

improve the signal-to-noise ratio. The most desirable is to increase the modulation depth. A, by increasing the modulator sensitivity. However, without materials with considerably larger

electro-optic coefficients, improvement in this component may be limited. Increasing the

optical carrier power helps, but only with the danger of photon-induced crystal damage.

Finally, narrowing the detection bandwidth brings substantial improvement for detecting CW

fields. If sufficiently narrow bandwidths to obtain good signal-to-noise ratios are

acceptable, low optical power and simple PIN diode receivers can produce excellent results

[24].

TRANSMI TTER S Desirable characteristics of a transmitter are that it be capable of

coupling adequate optical power into the uplink fiber, contribute relatively little noise to

the system, and have spectral purity sufficient to meet system bandwidth requirements.

Optimizing performance in one of these areas usually means compromising performance in

another. Often the input requirements of the modulator considerably restrict the options for

the source. In particular, modulators of the integrated-opt i cs type and some of the

polarization/analyzer designs require single-mode uplink fibers to match input-mode size and

prevent depolarization respectively. Only the output from solid-state or gas lasers can provide the necessary power densities in the few micrometer diameter core of single-mode

fibers. Lasers are also necessary to provide the spectral purity required to prevent material

dispersion from degrading system bandwidth if high-frequency operation over long fiber links

is necessary. If system bandwidth is limited by the modulator characteristics, and multimode

fiber can be used on both the uplink and downlink fibers, non-lasing, superluminous diodes may

provide a much lower system noise level with adequate optical carrier power.

The choice of the carrier wavelength depends on several system parameters. The most

important are the detector spectral response, the desired modulator sensitivity, and its

potential for optical damage. The first two requirements call for short wavelengths which, in

general, improve modulator sensitivity and for wavelengths between 600 to 900 nm to match the

spectral response of silicon photodiodes that are less noisy than the longer-wavelengthresponding GaAs-based devices. For LiNbO^-based modulators, photon damage has been seen to occur at wavelengths less than 1 . 4 ym [25]. Thus, in order to take advantage of shorterwavelength sensitivity, power levels in these modulators must be kept to a few uW and

operation at the HeNe wavelength of 633 nm should be avoided. Narrow spectral line widths are usually desirable to prevent chromatic dispersion in the downlink fiber from degrading system

96
bandwidth if high frequency and long fiber links are required. However, in most applications,

the fiber links are short enough that frequency response is limited by the modulator

characteristics instead of the fiber downlink. Spectral purity then becomes more of a concern

in regards to system noise.

The contribution of the transmitter to the system noise budget is both direct and

indirect. Electro-optic light sources are all directly affected with shot or quantum noise in

the photon-generation process. However, for a good laser that is operating above threshold,

the ratio of the noise power to the output power can be less than -IMO dB. In order to reach

these low levels of noise, stabilized single-mode operation is a must. This also avoids the

problem of mode hopping or mode partition noise that is a problem in multimode lasers.

Unfortunately, even single-mode lasers are easily destabilized and exhibit mode hopping if

optical feedback occurs due to reflections from fiber ends and other surfaces in the optical

system. This problem can be especially severe if modulated light is reflected back to the

laser. Every precaution needs to be taken in setting up a probe system to eliminate the

possible reflections and to make sure that those which do occur are not propagated back to the

laser.

Modal noise in multimode fibers and the less-severe polarization-mode noise in singlemode fibers are due to fluctuations in the energy distributed between the various modes when a very narrow spectral line (single-mode laser) is used. The effect is similar to the dancing

speckle pattern seen in laser light reflected from a rough surface. For this reason, a source with a broad spectral line is desirable if multimode fiber is used in a low-bandwidth system.

The characteristics of several candidate transmitter light sources are summarized in

Table 7.^. It is clear that the optimum system would use single-mode components throughout

the system. This should provide the greatest bandwidth, lowest noise, and highest

sensitivity. However, if system bandwidth requirements are low, a system designed to operate

with all multimode components and non-lasing superluminous diodes may provide adequate optical

carrier power and sufficiently low noise for considerably less expense than a single-mode

system.

FIBERS, CONNECTORS, AND COUPLERS As just mentioned, an electro-optic field probe can

be designed using either single-mode or multimode optical components. The highest performance

is obtained using an all-single-mode system, preferable operating at about 850-nm wavelength.

Single-mode fibers and connectors are presently available for such applications. However, the

availability of single-mode fiber for 850-nm operation in the future is of concern, since the

communication industry is moving to I.S'ym and 1.55-pm wavelength systems to increase repeater

spacing. Fibers designed for single-mode operation at these longer wavelengths will, in

general, be multimode at the shorter wavelength due to their larger core diameters and

resulting higher cutoff wavelengths [26]. Multimode fibers for 850-nm operation are readily

available because of their common use in local area networks. Both single-mode and multimode

fibers are available in multifiber cables, often in short sections left over from large

manufacturing runs. When using fibers in sensor applications, some care needs to be taken to

avoid sharp bends or stress which will induce severe birefringence.

Connectors are commercially available for both single-mode and multimode systems.

Multimode systems require less precision in aligning fiber ends and are therefore easier to

use. Connectors to the probe which are in the measurement field and near the antenna should

be nonmetallic in order to avoid field distortions. Plastic and ceramic units are available

to meet this need. Metallic connectors may be used at the transmitter/receiver unit.

Coupling light into fibers, especially single-mode fibers, is fairly difficult and is

sensitive to the system mechanical stability. Therefore, it is highly desirable to have

lasers with fiber pigtails and use all-fiber components to divide, switch and control the

light propagation. Fortunately, many of the components necessary for these operations are now available due to their development for the telecommunications industry. Their use also

increases the optical stability and considerably reduces the size and weight over that of a system using discrete open-beam-path components, such as beam splitter prisms, microscope

objectives, etc., and their usual mounting hardware. The continued rapid development of this

technology should make still better components available in the near future.

97
REFERENCES

[1] Bowman, R. R. Some recent developments in the characterization and measurement of

hazardous electromagnetic fields. Proc. Int. Symp. on Biolog. Effects and Health

Hazards of Microwave Rad; 1973 October; Warsaw, Poland. 217-227.

[2] Asian, E. Electromagnetic radiation survey meter. IEEE Trans. Instrum. Meas . IM'19:

368-372; 1970 November.

[3] Larsen, E. B. ; Andrews, J. R. ; Baldwin, E. E. Sensitive isotropic antenna with fiberoptic link to conventional receiver. Nat. Bur. Stand. (U.S.) Int. Rept. 75-819; 1976

September

.

[4] Kanda, M. A relatively short cylindrical broadband antenna with tapered resistive

loading for picosecond pulse measurements. IEEE Trans. Antennas Propag. AP-26(3): ^139- M47; 1978 May.

[5] Kanda, M. ; Ries, F. X. A broadband isotropic real-time electric-field sensor (BIRES)

using resistively loaded dipoles. IEEE Trans. Electromag. Compat . EMC-23(3): 122-132;

1981 August.

[6] Larsen, E. B. ; Ries, F. X. Design and calibration of the NBS isotropic electric-field

monitor (EFM-5), 0.2 to 1,000 MHz. Nat. Bur. Stand. (U.S.) Tech. Note 1033; 1981

March.

[7] Driver, L. D. ; Cruz, J. E. Development of the NBS isotropic magnetic-field meter (MFM10), 300 kHz to 100 MHz. IEEE Int. Symp. on EMC; 1982 September; Santa Clara, CA. M60-

467.

[8] Bensema, W. D. A noise spectrum measurement system using the fast Fourier transform.

IEEE Trans. Electromag. Compat. EMC-19(2): 37-43; 1977 May.

[9] Reeve, G. R. ; Synthesized isotropic pattern antennas for EM field measurements. IEEE

Int. symp. on EMC; August 1981; Boulder, CO. 36-42.

[10] Bensema, W. D. Broadband orthogonal array system microprocessor control and

computation. IEEE Int. Symp, on EMC; 1981 August; Boulder, CO. 43-46.

[11] Wyss, J. C.; Sheeran, S. A practical optical modulator and link for antennas. J. Lightwave Technology, LT-3(2): 31 6-320; April 1985.

[12] Bulmer , C. H. ; Hiser, S. C. Linear Ti: LiNbO modulators at 1 . 3 ym for electromagnetic

field sensing. Integrated Optical Circuit Engineering, SPIE, vol. 517; 1984.

[13] Kanda, M. An electromagnetic near-field sensor for simultaneous electric- and

magnetic-field measurements." IEEE Trans. Electromag. Compat. EMC-26(3): 102-110; 1984

August.

[14] King, R. W. P. The Theory of Linear Antennas. Cambridge, MA: Harvard University

Press; 1956. 184 and 487.

[15] Kanda, M. Analytical and numerical techniques for analyzing an electrically short

dipole with a nonlinear load. IEEE Trans. Antennas Propag. AP-28(1): 71-78; 1980

January.

[16] Yoshino, T. ; Ose, T. New approach to fiber-optic sensors using fiber Fabry-Perot

interferometers. paper WL6, CLEO Digest of Technical Papers, Optical Society of

America; 1982 April.

[17] Yao, S. K.; Findakly, T. ; Cordero-Iannarella, R. ; Thaniyavarn, S. ; Hayward, G. ; Chen,

B. Electromagnetic sensor using integrated optic channel waveguide modulator and

polarization preserving fiber. SPIE Proc, 412: 154-159; 1983.

[18] Yen, H. W. private communication, Hughes Research Laboratories, Malibu, CA.

98
[19] Wood, T. H. ; Burrus, C. A.; Miller, D. A. B. ; Chemia, D. S. ; Damen , C; Gossand, A. C;

Wiegmann, W. High-speed optical modulation with GaAs/GaAlAs quantum wells in a P-i-n

diode structure. Applied Physics Letter, 44: 16; 1984.

[20] Alferness, R. C. Waveguide electrooptic modulators. IEEE Trans. Microwave Theory and

Techniques, MTT-30(8): 1121-1137; 1982 August.

[21] Bulmer, C. H. ; Burns, W. K. Integrat ed-opt ic modulator for electromagnetic field

sensing. Paper THAAl , OSA/IEEE 3rd Int. Conf. on Optical Fiber Sensors; 1985 February;

San Diego, CA.

[22] Becker, R. A. Traveling-wave electro-optic modulator with maximum bandwidth-length

product. Applied Physics Letter, 45(11): 1168; 1984 1 December.

[23] Muoi, T. V. Receiver design for high-speed optical-fiber systems. J. Lightwave

Technology, LT-2(3): 243-267; 1984 June.

[24] Bulmer, C. H. ; Burns, W. K. Linear interf erometri c modulation in Ti: LiNbO . J.

Lightwave Technology, LT-2(4): 512-521; 1984 August.

[25] Halman, R. L. ; Busch, J. R. ; Verber, C. M. ; Wood, V. E. ; Hartman, N. F. ; Cressman, P. J. Z-axis propagation and the avoidance of optical damage in Lithium Niobate

waveguides. lEEE/OSA Seventh Topical Meeting on Integrated and Guided-Wave Optics;

1984 April; Kissimmee, FL.

[26] Jeunhomme, L. B. Single-Mode Fiber Optics, Principles and Applications. New York, NY:

Marcel Dekker, Inc.; 1983.

36

34

32

30

B> 28

03"V 2b

TJ

1—1 7A

0) •H

U^ 22

<U 20

4J

01u 18

T-l T)C ib

lA -

12

0.1 10 100

Frequency , MHz

1000 5000

Figure 7.1 Typical response of the NBS isotropic electricfield monitor (EFM-5)

99
DETECTOR PACKAGE

Ri c,

Di

R,= 10.3 Q. 1/2 W

D, -1N5711

EQUIPOTENTIAL

OUTER LOOP

FILTER NETWORK

R,= R^=R,= R,= 2.4kQ

RESISTANCE UNE

Figure 7.2 NBS magnetic-field meter (MFM-10)

'< -1

FIELD STRENGTH 0.2S5 A/m

Ail vaki*« normalized to IOMH2 value

rx^

100 0.3 1.0 10

Frequency, MHz

Figure 7.3 Normalized response curve of MFM-10

100
te'cOSlJ' ^;

LOAD

LOAD

Figure 7.4 A single loop with equal loads in a new probe for measuring

simultaneously the electric and magnetic fields.

10-3 ---} Theory •—• Experimental

10-4 re

9

o

10-5

10-e

|inl^(po«ItIv«)

ton LCpotltlvt] —7^

lml^(n«satlv«)

10 100

Fr«qu«ncy .MHz

1000

Figure 7.5 Magnetic-loop (Ij.) and electric-dipole (I^)

currents of a loop antenna with Z " 200 Ji.

101
Figure 7.6

^ experimental loop -model for measuring simultaneously

the electric and magnetic fields.

102
10

-4

fslOMHz

10

-5

c

0)u

u

3

o

10-V

-7

10 ' I

10-8 1 1 1 1 1 1 J

200 400 600 800 1000 1200 1400

ohms

Figure 7.7 Real parts of magnetic-loop and electricdipole currents of a loop antenna as a function of load impedance at 10 MHz.

103
ELEaRICALLY.

SHORT 01 POLE

Figure 7.8 Thevinin's equivalent circuit of a probediode system.

^^-_ NUMERICAL SOLUTION .._•— ANALTTICAL SOLUTIOM

_L

10' 10° 10' Frequency , Rz

10" 10' 10

10

Figure 7.9 Frequency response of a typical probe-diode system

with E^ =1 V/m. inc

104
Polarization Controller

Feedback

Antenna

Fiber Optic

Link

Figure 7.10 Block diagram showing basic concept of electro-optic

field probe.

105
Voltage

Figure 7.11 Characteristic operating mode for electro-optic modulator,

106
> I> oo ind d e«

CO >

09O

CD •HM

0) 4JU

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to uuo3

•ao«

.a

(0

E-t >

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ca4

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n i-t

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ino

00

w

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a.3e

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«M

r».

C« I > O 3 « — c e --

*^

m

a o ca

m iJ

w

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eoe«>«uu«u

eoe«

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01 eo e U

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4)u ca •a u o e

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41

u

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X ^4 3 m U

w* CL • P

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^-^ 1=

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>-> •o o O> 4J a X « ^4 w e e lu

41 aoO

4J ^14 4J m

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b o u 3 4) <U

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Q. eo

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<9 ^^ u •n

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107
Table 7.2 Principal Noise Sources for Optically Sensed Probes

Noise Sources

1 . Laser

Mitigation

use common mode rejection

detection scheme

Quantum noise

Mode partition noise

Optically feedback noise

2. Fiber

Modal noise

Modal polarization noise

3. Receiver

operate above threshold

stabilized, single mode

laser

a. use coupling optics to

reduce feed back

b. use Faraday isolator

a. carefully align and

match fibers and

couplers

b. use source with broad

spectral width

c. use single mode fiber

a. use low birefringent

fiber

b. use polarization

maintaining fiber

use low-noise amplifiers

carefully matched to

detector

Shot noise

Thermal noise in load

Dark current noise

limit carrier power

a. use avalanche

photodiode

b. cool electronics

insignificant for

this type system

108
Table 7.3 Signal and Noise Calculations for Typical PIN

and Avalanche (APD) Photodiodes

Signal current I so- MgrP A

Mean-square noise current I

- 2g^B [M^ F(M) erP ^ n o (shot noise)

M F(M) el^

d

(dark noise)

2kT

(thermal noise)

where

M

g

r

Bn F(M)

avalanche gain: 1 for PIN; 50 for APD

amplifier gain: 50 for PIN; 1 for APD

primary responsivity : =» 0.4 A/W for silicon photodiode at

A - 850 nm intensity of optical carrier: 50 yW

modulation depth of carrier by signal voltage: 0.01

bandwidth of detection system: 10 MHz

excess carrier noise factor: 1 for PIN; 4 for APD with M = 50

-19

charge of electron: 1.6(10) C

detector dark current before gain : 200 pA

Boltzmann constant: 1.38(10)

temperature: 300 "K

load resistance: 50 Q

-23

J/^K

PIN APD

I^ - 1 .6(10)"" (shot)

n

+ 1 .6(10)"^^ (dark)

+ 8.25(10)"^ (thermal)

- 8.3(10)"^ A^

(10)-^° A^

S/N 1 .2(10)

-2

. 6.4(10)"''^ (shot)

+ 6.4(10)'^^ (dark)

+ 3.3(10)'^^ (thermal)

6.7(10)

-1

(10)-'° A^

1.5 (just detectable)

109
Table 7.^ Typical Characteristics of Several Light Sources

for Electro-Optic Sensors

Source

Wavelength

(nm)

Spectral

line width

(nm)

Single

Multi

Mode (SM)/

Mode (MM)

Output

Power (mW)

Light

emitting

diodes

Ga Al As

Ga As

In Ga As

780-850

940

1060

• 40

- 40

50

NA

NA

NA

2-10

2-6

0.2

Laser

diodes

Ga Al As

Ga Al As

In Ga As P

780-850

780-850

1300

2 - 3

0.005 - 2

3

- 4

MM

SM

MM/SM

2-10

2-10

1 - 10

Gas lasers

He Ne

He Ne

632.8

1152

5 MHz*

5 MHz*

SM

SM

0.5 - 10

0.05 - 1

* 5 MHz corresponds to 6.5(10)

-15-6 m - 6.5(10) nm.

110
Chapter 8. REVERBERATING CHAMBERS

The limitation on operating frequency for TEM cells as described in Chapter 3 could be a drawback for certain applications. Furthermore, since the polarization of the field generated

inside a TEM cell is fixed, the radiated emission and susceptibility tests for an EUT using

TEM cells require physical rotations (or different orientations) as discussed. This

requirement of EUT rotation could be another inconvenient aspect.

A relatively new EMC/EMI measurement technique, which does not require EUT rotations, is

to utilize reverberating chambers to generate an average uniformly homogeneous and isotropic

field within a local region inside a metal enclosure [1]-[9]. A reverberating chamber is a shielded room with a rotating stirrer to mix the field generated by a transmitting antenna.

The polarization of this homogeneous and isotropic field is randomly varying. It is precisely

because of this special property that the physical rotation of test objects can be avoided.

The homogeneous and isotropic field is achieved by rotating an irregularly shaped mode stirrer

or tuner, either continuously or in steps [7]. Naturally, the associated boundary conditions

are changing with respect to time so that the possible modes existing simultaneously inside a given shielded metal chamber (cavity) are perturbed accordingly.

Two analytical approaches for treating this type of EM field problem and for providing

basic knowledge for design purposes are possible. One involves the direct solution of

Maxwell's equations with time-varying boundary conditions. A formal solution using this

direct approach is rather difficult to obtain. In the other approach, suitable linear

combinations of basic eigenmodes of the unperturbed cavity (without mode stirrer or tuner)

with time-dependent expansion coefficients are taken to represent the field and to satisfy

approximately the boundary condition on the surface of the rotating mode stirrer or tuner [7].

The main advantage of this latter approach is that the unperturbed eigenfrequencies and

eigenmodes are much easier to calculate, and the problem can be reduced to a more familiar one under special conditions. A necessary condition for the validity of this method is, however,

that the total number of eigenmodes which can exist inside a chamber be large for a specified

frequency and chamber size. Thus, the measurement technique using reverberating chambers is

good for very high frequency application, and may serve as a powerful supplementary tool to

TEM cells. Typical frequencies of operation are from a few hundred megahertz to 20 GHz or above.

The reverberating chamber is also capable of providing a very efficient conversion of source power to high-strength fields inside a shielded enclosure for performing EMC/EMI tests

of large equipment and whole systems. The limiting factors are that users may have

difficulties interpreting the measurement results taken inside the chamber and relating them

to actual operating conditions, and that polarization properties are not preserved for

characterizing an EUT.

8.1 Some Design Considerations

As expected, the total possible number of modes, N(f), inside an unperturbed, lossless,

rectangular chamber of dimensions (a x b x d) increases in steps with frequency. A smooth

approximation to N(f) may be given by [7], [8]:

3 N (f ) = 1^ abd -^ - (a + b + d) - + ^ , (8.1)

s 3 C 2 c

where abd in cubic meters represents the chamber volume, f the operating frequency in hertz,

and c the speed of wave propagation in the chamber (usually air) in meters per second.

Note that the first term in (8.1) is identical to Weyl's formula [7] derived originally

for the same problem by a different approach, and is proportional to the chamber volume and

the third power of frequency. The second term may be recognized as the edge term, which is

proportional to the sum of the linear dimensions of the chamber and modifies Weyl's result;

this term is especially significant in the lower frequency range. The inner surface area of

the chamber, S = 2(ab + bd + da), is not involved in (8.1). An example is given in Fig. 8.1

showing N(f) as Curve 1, N (f) as Curve 2, and Weyl's formula as Curve 3 for the NBS chamber.

Note that the dimensions of the NBS chamber, 2.74m x 3.05m x 4. 57m, are all unequal. Designs

of chambers having two or three sides equal but with the same volume will increase mode

degeneracy, thus decreasing the total number of distinct modes with respect to a given

operating frequency. Under this condition, wider steps in Curve 1 than those shown in Fig.

Ill
8.1 will be observed, even though the smooth approximation (Curve 2) remains almost unchanged.

To demonstrate this point, specific examples are shown in Fig. 8.2 for a square-based chamber

(2.17m X 4.19m x U.19m) and in Fig. 8.3 for a cubic chamber (3.37m x 3.37m x3.37m).

While the total number of eigenmodes inside an unperturbed chamber is an important design

criterion, another equally important factor to consider is the mode density function, dN/df,

which represents the change in the number of modes in a given frequency interval. Of course,

the exact shape of dN/df involves impulse functions as it is the derivative of step functions.

An alternative quantity for exhibiting this property is to examine

AN = /^^ f df . (8.2)

which represents the increase or decrease in mode number within a frequency interval of Af.

Results of AN when Af = 1 MHz are presented in Fig. 8.4 for the NBS chamber, in Fig. 8.5

for the square-based chamber, and in Fig. 8.6 for the cubic chamber, all with the same volume.

Clearly, the uniformity of mode distribution in the frequency interval of 1 MHz is good for

the NBS chamber, fair for the square-based chamber, and poor for the cubic chamber, where no change in mode number is possible in a relatively larger portion of frequency sub-bands. The

general design criteria for a reverberating chamber are then to make the volume as large as possible and the ratio of squares of linear dimensions as nonrational as possible.

A third design criterion, namely, the quality factor (Q), should also be considered to

characterize a chamber when it is made of lossy material. Since there are so many modes

existing in an unperturbed chamber with each mode carrying its own Q value [7] , [8] , [1 0], it is

not trivial to define a quality factor for the chamber as a whole. One method of defining a composite quality factor for an unloaded chamber (without an EUT in it) within a specified

frequency range gives the following simple, approximate result [7]:

Q = I

[V/S6^] . :r^ -. -. (8.3)

8k ^a b d ^

where V denotes the chamber volume (abd) in cubic meters, and 6 the skin depth in meters of

the material of which the chamber is made.

The physical meaning of (8.3) may be interpreted by comparing it with the individual Qvalues of all the modes in the form of a cumulative distribution. Since V/(S6 ) is a common

s factor whether the composite quality factor defined above or the quality factor for individual

modes is considered, it is more convenient to present the results in terms of 1 /Q values

normalized with respect to S6 /V. Thus, the variable used herein is

a = i (V/S6 ) . (8.4)

Q s

Examples of the cumulative distribution of a for the NBS chamber are given, respectively, in

Figs. 8.7-8.9 for three different frequency bands. For the frequency band of 1 80 to 200 MHz

in Fig, 8.7, the total number of modes existing in this band of 20 MHz is 69, with each mode

having its own Q-value. The probability of having a high upper-bound value of a ^ 0.80 (or a lower bound for Q) is almost 100 percent, and that for a low bound of a ^ 0.48 (or a high

value for Q) is only about 10 percent. This implies that almost all of the 69 modes in this

particular frequency band have a i 0.80. The arithmetic mean of 0.623 and the standard

deviation of 0.090 are also indicated in the figure. The probability of having a ^ 0.623

(arithmetic mean) is, of course, 50 percent, meaning that at least one half of the 69 modes

have a i 0.623.

For the case presented in Fig. 8.8, where the frequency band is from 330 to 350 MHz, also

of a bandwidth of 20 MHz, there are 261 modes, an increase in number of modes relative to that

in Fig. 8.7. This is because of higher frequency. A similar interpretation of the a-values

(or Q-values) carried by these modes in terms of probability applies. Note that there are now

a small number of modes (low probability) carrying a value of a as low as 0.43 (high Q). The

arithmetic mean and standard deviation are, respectively, 0.630 (higher than the corresponding

value in Fig. 8.7) and 0.085 (lower than the corresponding value in Fig. 8.7). A higher value

of arithmetic mean implies that one half or more of the 261 modes carry a higher value of a

112
(lower Q) compared to the frequency band considered in Fig. 8.7. A decrease in standard

deviation reveals that a greater number of modes have an a value closer to the arithmetic

mean.

If we consider a still higher frequency range 480 to 500 MHz, such as that illustrated in

Fig. 8.9, 534 possible modes will exist in the same bandwidth of 20 MHz. The arithmetic mean

increases further to 0.646 while the standard deviation decreases further to 0.074, indicating

that a greater number of modes will have still higher a values near the arithmetic mean. This

general tendency, Increasing in arithmetic mean and decreasing in standard deviation with

increased frequency, yields a limiting mean of a ^ 0.667 with a 50-percent probability, which

also agrees precisely with the limiting value for the composite a derived from (8.3) and

(8.4).

Thus, even though there are a large number of possible modes existing in a specified

operating frequency band for a reverberating chamber, with each mode carrying its own value of

a or Q, the probability that a i 2/3 (or Q a 1.5V/S6„) is 50 percent. This fact implies that

one half of the modes have a values less than 2/3. Preliminary estimation of a quality factor

to characterize the reverberating chamber as a whole, based on the simple expression Q in

(8.3) » for the purpose of predicting the field-strength level to be generated in the test

zone, is indeed very useful. The composite quality factor so estimated is considered as the

upper-bound value because it does not take into account losses other than that due to wall

conductivity.

Since compromises between low conductor losses (high Q) and broad model coverage (low Q) are almost always necessary in the practical design of reverberating chamber, the results for

AN in (8.2) and Q in (8.3) will be found convenient and helpful.

8.2 Recent Measurement Results

The basic measurement system used at NBS is shown in Fig. 8.10. The test field is

established by means of one or more RF sources feeding one or more transmitting antennas

placed inside the chamber. We use the log-periodic antenna (0.2 to 1.0 GHz), corrugated horn

(1.0 to 4.0 GHz), or double-ridged circular horn (4.0 to 18 GHz) as the transmitting antenna.

Modes excited inside the chamber by the transmitting antenna are then tuned or stirred by

rotating one or more field-perturbing devices referred to as "Paddlewheel tuners". The tuners are typically electrically large metal blades or irregularly shaped structures that are mounted on the enclosure walls or ceiling driven by electrical motors. The reason for using

the irregular shape is to help scatter the field more evenly in all directions. Figure 8.11

shows the NBS tuner. The field generated at the test zone is then measured by a calibrated

probe as a function of frequency and tuner position. Note that there is another receiving

antenna placed inside the chamber to measure the maximum and minimum received powers (see Fig.

8.10). The long-wire antenna (0.2 to 1.0 GHz), corrugated horn (1.0 to 4.0 GHz), or double

ridged circular horn (4.0 to 18 GHz) is employed as this receiving antenna. The measured data

of these received powers are useful for assessing the effectiveness of the tuner, the relative

uniformity of the field, and the quality factor of the chamber.

In the mode-stirred case [1],[6], the tuner rotates continuously, thus changing the

chamber boundary conditions, the voltage standing-wave ratio (VSWR) of the transmitting

antenna, the net input power to the chamber, and the field polarization and strength. The

resultant field is then recorded by detectors with sampling rates much faster than the tuner's

rotating speed, and statistically processed. In the mode-tuned case [9], [11], the tuner

position is stepped at discrete intervals. Measurements of the net power delivered to the

transmitting antenna, power received by the reference antenna, and perhaps also the monitor

response of the EUT being tested are taken for each tuner position and frequency. This mode

of operation makes possible corrections for the variations in field strength due to changes in

the VSWR. An example of the transmitting antenna VSWR variation in the NBS chamber is

presented in Fig. 8.12, giving the maximum, average, and minimum VSWRs as a function of

frequency, obtained by rotating the tuner through a complete revolution with 200 steps (1.8°

increments). Note the large variation at the lower-frequency end. Failure to correct for

this change could result in errors in determining the test field level [12]. At higher

frequencies above approximately 2 GHz, variations in VSWR become much more reasonable.

To evaluate the tuner effectiveness, the ratio of maximum-to-minimum received powers for

a complete revolution at 200 steps is shown in Fig. 8.13 as a function of frequency, showing

an average of 30 dB. The net input power to the transmitting antenna has been maintained

113
constant for this measurement. If the tuner were not as effective, the maxiraum-to-miniraum

received power ratio would be very low. This fact can be verified by using a much smaller

size of tuner with a rather regular shape.

Another important feature characterizing the chamber is the measured difference between

the net input power to the transmitting antenna and the power available at the reference

antenna terminals. This difference represents the chamber loss, which may be used to estimate

the actual mean quality factor of the chamber. This information, of course, is important for

considering the power requirement and broad modal coverage. The loss characteristics for the

NBS chamber, both the average and minimum losses obtained by rotating the tuner by one complete revolution in 200 steps, are shown in Fig. 8.14. The smooth solid curves represent

estimated fits from the measured data. Based on the measured minimum loss data, the actual mean chamber quality factor (Q') may be estimated by [2]:

Q' = ^e/ -^ (P^/PJ . (8.5)

where P is the minimum power available at the reference antenna terminals, P^ is the net

r

^

t power delivered to the transmitting antenna in the same unit as that of P , V is the chamber

volume in cubic meters, and X is the operating wavelength in meters.

The result of (8.5) is plotted in Fig. 8.15 as Curve (a). For comparison purposes, the

composite quality factor (Q) as computed by (8.3) is also shown in the same figure as Curve

(b). The ratio of Q/Q' is presented in Fig. 8.16, showing that Q' approaches closer to Q for

higher frequencies where more modes are available and the field strength inside the chamber is

relatively more uniform.

The maximum and average electric-field strengths generated inside the chamber, as adjusted to 1 -W net input power, are presented in Fig. 8.17 as a function of frequency. The

field strengths were measured using an electric-field probe (1-cm dipole) placed at the center

of the test zone, with 200 tuner positions. The probe was rotated through three orthogonal

orientations aligned with the chamber axes. The magnitude of the total electric field was taken as the square root of the sum of the squared values of the three components. The probe

response was calibrated in a planar field environment at frequencies up to 2.4 GHz. The field

strengths at frequencies above 2.4 GHz were also determined by using the same probe response.

An increase in field strength at approximately 15 GHz as shown in Fig. 8.17 is believed

due to the probe resonant characteristic rather than due to the chamber. It can be corrected,

if necessary, by recalibrating the probe at 15 GHz. Also included in Fig. 8.17 are the field

strengths calculated from the power P received by the reference antenna in accordance with

the following:

E = -^ (30P

)'^'' V/m . (8.6)

c A r

The above expression was derived based on the assumption of far-field environment. Obviously,

such condition does not exist inside a multimode chamber. The expression has been used

frequently within the EMC community even though it has received only marginal justification

[1]. The results presented in Fig. 8.17, however, demonstrate good agreement between the

direct measurements and that derived from (8.6). To increase further our own confidence, we

also independently measure the maximum, average, and minimum magnetic-field strengths with a magnetic-field probe (a loop antenna of 1-cm diameter) in a similar manner. The corresponding

ratios of the electric field to the magnetic field may be loosely referred to as wave

impedances. Results for the wave impedance are given in Fig. 8.18, showing wide variations,

as expected, with frequency. However, the average wave impedance at frequencies above 200 MHz

(the lower frequency limit recommended for the NBS chamber) is approximately 120it. This serves as an additional check to the validity of (8.6). Thus, it appears that (8.6) can be

used as a preliminary means to determine the level of the test field.

It is important to note that the reference antenna type is not significant, except that

it is desirable to have its impedance reasonably matched to the power detector, especially for

high frequencies when more modes are available in the chamber, so that its average VSWR

behaves well. Hence, antennas within their designed frequency bands may be used as a reference. Their gain characteristics are rather unimportant.

114
After a reverberating chamber is designed and its basic characteristics determined as outlined above, the radiated susceptibility measurements can be performed by placing the EUT

at the test zone, routing its control cables to monitors outside the chamber, and preventing

leakage to the exterior environment by using proper filtering processes, if needed. A test

field has, of course, to be generated first by supplying power to the transmitting antenna.

The desired field level may be established, as required, by gradually increasing the input

power. The EUT operation is monitored for malfunction while the tuner/stirrer is rotated

through a complete revolution. The rotating rate should be slow enough to allow sufficient

time for the EUT to respond to the change in test-field level. If a malfunction of EUT is

observed for a particular tuner/stirrer position, the tuner/stirrer should be stopped and the

net power to the transmitting antenna should be reduced until the malfunction ceases. The

test results are then recorded and the test proceeds to the next frequency, etc., until the

test is completed.

Radiated emission measurements are performed in a fashion reciprocal to susceptibility

measurement or by using a substitution technique. In this case, the transmitting antenna is

removed from the chamber because the EUT itself is the radiator. The power received by the

reference antenna may be measured and used to determine the total power radiated by the EUT

with the aid of the chamber quality factor (8.5), or the equivalent radiated far field by

(8.6). Alternatively, the power radiated by the EUT may also be determined by measuring the

equivalent power applied to a transmitting antenna substituting for the EUT, provided, of

course, the same power is received by the reference antenna.

REFERENCES

[1] Bean, J. L. ; Hall, R. A. Electromagnetic susceptibility measurements using a modestirred chamber. IEEE Int. Symp. on EMC; June 1978; Atlanta, GA.

[2] Corona, P. Electromagnetic reverberating enclosures: behavior and applications. Alta

Frequ., M9, 54-158; 1980.

[3] Corona, P.; Latmiral, G.; Paolini, E. ; Piccioli, L. Performance of a reverberating

enclosure for power measurements in the microwave range. Proc. second Symp. Tech.

Exhibition on EMC; 1977; Montreux, Switzerland. il19-423.

[4] Corona, P.; DeBonitatibus , A.; Ferrara, G. ; Paolini, E. Electromagnetic enclosures

behavior — comparison data analysis. Proc. third Symp. Tech. Exhibition on EMC;

1979; Zurich, Switzerland. 231-235.

[5] Corona, P.; Latmiral, G.; Paolini, E. Performance and analysis of a reverberating

enclosure with variable geometry. IEEE Trans. Electromag. Compat . EMC-22(1): 2-5;

1980 February.

[6] Corona, P.; Latmiral, G.; Paolini, E. ; Piccioli, L. Use of a reverberating enclosure

for measurements of radiated power in the microwave range. IEEE Trans. Electromag.

Compat. EMC-18(2): 54-59; 1976 May.

[7] Liu, B. H. ; Chang, D. C; Ma, M. T. Eigenmodes and the composite quality factor of a reverberating chamber. Nat. Bur. Stand. (U.S.) Tech. Note 1066; 1983 August.

[8] Liu, B. H. ; Chang, D. C; Ma, M. T. Design consideration of reverberating chambers for

electromagnetic interference measurements. IEEE Int. Symp. on EMC; 1983 August.

Arlington, VA.

[9] Mendez, H. A. A new approach to electromagnetic field-strength measurements in

shielded enclosures. Wescon Record; Los Angeles, CA; 1968 August.

[10] Harrington, R. F. Time-Harmonic Electromagnetic Fields, New York, NY: McGraw-Hill;

1961 .

[11] Roe, J. M. An improved technological basis for radiated susceptibility and emission

specifications. IEEE Int. Symp. on EMC; 1978 June; Atlanta, GA.

[12] Crawford, M. L. Evaluation of a reverberating chamber facility for performing EM

radiated field susceptibility measurements. Nat. Bur. Stand. (U.S.) Int. Rept . 81-

1638; 1981 February.

115
NBS chamber

« a = 2 . 74m

b = 3.05m

40 50 60 70 80 90 100 110 120 130 140 150 160 170 180 190 200 210 220

f, MHz

Figure 8.1 Total number of modes as a function of operating

frequency for the NBS chamber.

116
120 -•

110 -•

100 -

90 --

80 -

70 -.

60 --

50 ..

40 --

30 -.

20 -

10 .-

.-

a = 2.17Tn

square-based chamber \ b = 4.19ni

d = 4.19in

(1 , (2) , and (3) have the same meanings as those in figure 19.

-P^ '.:;;^

I 1 • >

-\ 1 1- -\ 1- H 1

\ 1- 40 50 60 70 80 90 100 110 120 150 140 150 160 170 180 190 200 210 220

f, MHz

Figure 8.2 Total number of modes as a function of operating

frequency for a square-based chamber whose volume

is the same as that of NBS chamber.

117
120

110

100

90

80

70

60

50

40

30

20

10

cubic chamber : a = b = d = 3.37m

1,2, and 3 have the same meanings as those in figure 19.

40 50 60 70 80 90 100 110 120 130 140 150 160 170 180 190 200 210 220

f. MHz

Figure 8.3 Total number of modes as a function of operating

frequency for a cubic chamber whose volume is the same as that of NBS chamber.

118
a = 2.7Am

NBS chamber <* b = 3.05m

d = 4.57m

AM

24 -

20

16

12 +

6

4 -

,N- ( ^

J df

AN - I — df with Af - 1 HHz

At

UL n rjl rinnnn MJIii nn . iln ll ^rlHP r^ II . 1

140 150 160 170 180 190 200 210 220 230

f, MHz

Figure 8.4 An illustration of mode degeneracy (NBS chamber)

AN square-based chamber: a = 2.17m, b == d = 4.19m

24 -

20 -

AM - r ^ df with Af - 1 MHz

J df

Af

16 - -

12

fl -

•

4

-

- —L-l—»- n n A

_JJ—^J

LL,—11^ 1 ' ' 1

' -L-H—

"L n

1

L|

140 150 160 170 180 190 200 210 220 230

f, MHz

Figure 8.5 An illustration of mode degeneracy (square-based chamber)

119
/\H

24

20 t

16

12 -

8

-

cubic chamber: a = b = d = 3.37m

kN - r dN

df

df with At . t MH: Af

' " I a

140 150 160 170

i Lin LLj

180 190

f, MHz

200

t—

210 220 230

Figure 8.6 An illustration of mode degeneracy (cubic chamber)

JO

COo

u

r a = 2.7'!im NBS chamber < b = 3.05m

lOOJi - •- 1 d = 4.57m

J

90 -

80 -

180 ^ f « 200 MHz (69 modes)

, fj

/

70

60

-

- /

^0% - - f

y

arithmetic mean = 0,623

40 - -

, /

stand, deviation = 0.090

30 -

- 20 - . H

10 -

h

, r^

J-. , ^ 1 1 , ^_ a

0.40 0.45 0.50 0.55 0.60 0.65 0.70 0.75 0.80

Figure 8.7 Cumulative distribution curve of the normalized 1/Q values

in the 180-r-lHz to 200-MHz frequency band for the NBS chamber.

120
ij

(U U) U3

(U E 3 nj

t-^ J=

<0 (J >

CO

C CQ ^^ z

I—

(U 13 J=

(U JJ N

•Hu

1—

o

CO <4-l ^ T) O c c to ja

(U A >^

4-1

ac

IM (U O 3

cr

0) (U > M

l-l (« 3a N c g o 1 •H c

4-1

m 3 en

U3

•H o

>^ 4.1 4J

CON

•H •X. T3 s:

(U 1 c ? <r\ •H en

4-1 CO 0) nH J= 3 jj E3 C

CJ •H

CO00

0)

l-l 3

oc

A3-pxTq«qoja

121
o

000)os

•a- enm oo

V0Ooooo u

(U

CO J3

(U E 3 CO i-t <— CO u > w

c- aaz

t-t

(U •o J^

(U 4J N

•H

u

1—t o

CO VM E

>-i T3 o C c CO XI

(V f >N

JJ oc

(4-1 0) o 3

u-

(U (U > u

ll U-l 3u N c 2 o 1 •H o u c 3 U1 J3

•H o

;^ 4J

cn

N

a; o > CX) •H >T

4J

(0 0) <—(

J= 3 J-l E3 c O •H

<r>• 00

<uu3

00 oo o o

\0 o o oiTv ^r i«>

O

^^TiTqsqojd

122
00aaou3

CQ

(8O

(0 •M

CQ>«

CQ

(0

b>3

COo

ftoc

(0 •o

(]} c3

<0o6

•o

(0os

C9

*^Os

eiu

to 9u3

(SO

123
Fieure 11 Photograph of tuner inside NBS reverberating chamber.

124
V5NR OF TRANSMITTING RNTENNR

i i I 1 I 1 T"

FREQUENCY (GHz)

Figure 8.12 Variations in VSWR of transmitting antennas placed

inside the NBS reverberating chamber.

Log-periodic: 0.2 - 1.0 GHz,

Corrugated horn: 1.0 - 4.0 GHz,

Double-ridged circular horn: 4.0 - 18.0 GHz.

125
Frequency , GHz

Figure 8.13 Maximum to minimum received power ratio as a function of

frequency for the NBS chamber and tuner.

43

ca!0

n

o

•J

T—I I I I I I I I I I I I

20

Frequency , GHz

Figure 8.14 Loss characteristics for the NBS chamber (empty)

with the smooth solid curves representing estimated

fits from the measured data.

126
m

u

01 I

o

9 . E+5 -

7.E+5

5. E+5

3. E+5

J. E+5

Frequency, GHz

Figure 8.15 Quality factor of the NBS chamber: (a) experimental mean Q' obtained from (8.5) based on j^ieasured received

power, and (b) theoretical composite Q as compared

from (8.3).

c

43 -

35 -

25 -

13

.2

I I I I I I I J I I I I I I 20

Frequency, GHz

-Aj Figure 8.16 Ratio of theoretical composite Q to experimental mean Q '

127
>

pa

H

^3

35

25

13

5 -

Mill 1 1—I I I I I

hff^XIMUM riELD

+++ riELD TROM CflLIBRHTED 1 CM DIPOLE

oooo FIELD CflLCULflTED FROM RECIEVE flNTENNfl

.2

J \ I I I I I I J I I II I20

Frequency, GHz

Figure 8.17 Maximum and average electric-field strengths generated

inside the NBS chamber (empty, mode-tuned).

2eee

1802

isaa

I4ee K

§

•fe isro

01c 1B00

a

•ou

Q.E

•H eee

«

>3 BBQ

468 -

286 -

T—'—I—'—I—•—J—'—

r

1—'—

r

Average

Uinlmvun

l__j I I L Jl.

120TT

56 IBB 13B 288 258 388 358 488 458 SBB 55B

Frequency, GHz

Figure 8.18 Wave impedance of the NBS chamber.

128
Chapter 9. CONTEMPORARY EMC TOPICS

9. a MEASUREMENTS IN, AND OF, COMPLICATED ELECTROMAGNETIC ENVIRONMENTS

It is quite appropriate that the subject of complex EM environments occurs in the

Contemporary EMC Topics chapter of this short course, for it will present more problems than

solutions. The field is still in its infancy, and many of the problems are not yet even welldefined, much less solved. Nevertheless, it is a very important area in EMI/EMC, and one that

cannot be ignored.

The standard methods of electromagnetics tend to assume and exploit various simplifying

features such as single frequency, single source, plane-wave fields, simple geometry, etc.

There are a number of good, valid reasons for making such assumptions: a) it may be

pedagogically desirable to keep examples free of non-essential complications; b) simplifying

assumptions may be necessary in order to solve a problem at all (Some problems are just

intractably complex.); c) for linear problems, the multiple-frequency solution can be

constructed by the superposition of single-frequency solutions; d) in design problems, for

example, one is free to choose simple geometries if they permit the design of the device or system desired. However, complicated problems do still exist. The real world is full of EM

environments with multiple (often unknown) sources, both primary sources and scatterers, with

multiple frequencies and nonsinusoidal waveforms, complicated asymmetric geometries, near

fields, and nonlinear devices or materials. These complex environments may involve sensitive

electronics, ordnance, communication systems, and people. The problem is to develop methods

of measuring and characterizing an EM environment so that one knows whether it will affect

whatever goes into it. The question of just what EM characteristic is important (maximum

electric field, average total power density, fraction of the time the field exceeds some threshold value, etc.) depends on the device or system, and is an interesting problem in

itself. We will concentrate on how and what does one measure in a complex EM environment.

Two separate questions will be addressed: how do probes react in a complex environment,

and how can one obtain useful information from relatively few measurements. In treating probe

response, we first consider actual errors in a meter for measuring the electric field and then

the errors inherent in trying to measure total power density using a probe which is sensitive

only to either the electric or the magnetic field, but not both. This problem will be

discussed in Section 9.a.1. Section 9. a. 2 is devoted to measuring characteristics of complex

environments. The primary emphasis is on a statistical approach, but we also briefly mention

other approaches currently under development.

9.a.1 An Electric-Field Meter (EFM) in a Complex Environment

A. Meter Errors This section will follow closely the treatment of [1], to which the

reader is referred for details and derivations. We consider first the errors in the electric

field actually measured. EFM's are typically calibrated using single-frequency, single-source

standard fields, and their response to more complex fields can be different than for the

calibrating fields. The type of EFM we have analyzed is a common (see e.g., [2])

configuration exemplified by, but not restricted to, the NBS EFM-5 [3]. (When specific values

of meter parameters were needed, those of the EFM-5 were used. The qualitative results,

however, apply to any meter sharing the same design.) The meter consists of a short dipole

antenna with a diode detector as a probe, with a transmission line connecting it to the box

containing the metering electronics. For an isotropic probe, three orthogonal dipole antennas

are used, each with its own independent transmission line within the cable joining probe and

box. Details are described in Chapter 7.

Because the dipole antenna is short compared to the wavelengths of the radiation being

measured, the electric field does not vary significantly over the volume of the probe, so that

it effectively is sensitive to the electric field at a single point as a function of time, E

(x = 0, t). An important consequence follows immediately: for meter errors the number of sources is immaterial, and all that matters is the temporal waveform of the electric field at

the probe position. Only if the field's time dependence is a single sinusoid is the probe

operating under the calibrating conditions. A corollary result is that the presence of

reflections does not introduce new multiple-source and frequency errors since they have the same frequency as the field from the primary source and combine with it to yield a net E (x = 0, t) which is still a single sinusoid in time.

When the incident field at the probe position does not have a single sinusoid for its

temporal waveform, errors arise from a collusion of the nonlinear response of the diode

129
detector and the filtering effect of the transmission line. The antenna plus diode plus

transmission line can be represented by the equivalent circuit of Fig. 9.a.l. The filter line

is represented only by a black box which is assumed to transmit dc , filter RF, and have little

effect on the effective load of the antenna. The input voltage v.(t) is proportional to the

electric field at the probe position; and an expression for the output voltage v (t) can be

obtained in terms of the input voltage and three parameters of the antenna-diode combination.

The three parameters are an effecti^

which for the EFM--5 have the values

-1 -1 -1 The three parameters are an effective time constant (a ) and two voltage scales (b , a ),

a = 33.8 kHz, b = ^1.22 V \ a = 38 v"''

.

(9.a.1)

Provided the input signal is periodic, with period (p) short compared to a (ap << 1 ) , the

voltage across the diode can be written

v^(t) = b a'^ v.(t) - a''' in {p'^ f e^ "^i^^'^dt'}

(9. a. 2)

V. (t) = v. (t) - <v.>

Note that the argument of the logarithm is time-independent, which is not true at lower

frequencies. The effect of tY resulting in an output signal offrequencies. The effect of the filter line then is to remove the RF component of v (t),

V (t) = - a'^ iln 1/ e^ ^^^' = ^P^dx} . (9.a.3)

In the limiting cases of small or large incident fields (blv.l << 1 and b(v.) >> 1

' I'max 1 max

respectively), the output voltage assumes the simple forms

V = - b^(2a)'^ <v.^> for small Ie.I,

o 1 ' 1

'

= - b a V. for large |E. I

,

(9.a.M)

where V. is the peak value of v.(t). These are the usual square-law and linear regimes; the

crucial point is that the square-law and linear dependences are on different variables. For

small incident fields, the signal reaching the meter is directly proportional to the average

2 of (E. - <E.>) , and completely insensitive to the peak value, whereas for large incident

fields the opposite is true. Therefore, only the average reading need be correct for small

fields and only the peak reading for large fields (both assuming v. = v.). For intermediatemagnitude incident fields, neither peak nor average reading need be correct. The magnitude of

the meter error depends on the actual waveform, the field intensity, and whether peak or average is measured.

In order to compute the error for a given E.(t) [and hence v.(t)], we first use (9. a. 3)

to calculate the resulting output voltage v . We then find the value of V such that an input

voltage of V cosoot would produce the same v . Since the meter was calibrated with a single

o sinusoid, it associates V cos ojt with v , and it will read V , (meter) = V and V (meter) = o pk avg

2 1/2 2 (V /2) . The errors in peak and average readings for E are therefore given by

A , = 10 log (v^ /V^) pk * 1 max

A =10 log (<v^>/0.5V^)

avg 1

(9. a. 5)

130
The errors calculated in this manner for a few representative waveforms are shown in

2 Figs. & 9. a. 2 -9. a. 4. In each case, A , the dB error in the average E , is plotted as a avg ^ ' ^ function of incident field strength. The A , value is related to A , as given in the figure

pk avg ^ captions. For a waveform like a narrow pulse, the error in the measured peak or average

strength can be as large as 10 dB, but more typical values are on the order of 2 dB. These

results are frequency-independent, provided ap << 1 (f > 1 MHz for the EFM'5).

We also investigated the case of various numbers of randomly distributed sources with

different strengths and frequencies, finding that the average meter error was around 2 to 'i dB, but that the maximum possible errors were much larger.

A few qualifications and clarifications of these results should be noted. The first

point is that our analysis is only valid for incident fields with periods which are short

relative to a certain characteristic time scale of the antenna plus diode. For longer-period

signals including those with only high-frequency Fourier components such as AM radio, our analysis does not apply. In addition, even for short periods, it is quite possible with the

EFM configuration considered here to choose the meter parameters such that one is always

effectively in the large-field domain or always in the small-field region; and therefore it is

possible for the meter to measure accurately eit her the peak field or the average field

squared, even for complex, multif requency fields. It is not, however, possible to measure

both peak and average with a single probe of this configuration, and meters which try to do so will err on one or the other (or both) at any given field strength.

B. Energy-Density Inference Errors Meters which only measure either the electric or the magnetic field cannot measure the true total EM energy density. That requires

simultaneous measurement of both electric and magnetic fields. In order to infer a total

energy density from a measurement of either the electric- or magnetic-field strength, some relationship between electric and magnetic energy densities must be assumed. The assumption

usually made is that the two are equal, as is true for a single plane wave. This section

considers the errors involved in that assumption.

We define

A(E/H) = 10 log (2u„/u^ _2J1]2 J = 10 log ^ II ^ , (9. a. 6)

^ ^°^ |E|2 ^ (y/e)|H|^

where u„ and u, . are the electric-field and total energy densities respectively. It is clear

b tot

that this "error" is a property of the environment and not of the meter used; in fact we shall assume that the electric field is known exactly. Thus, A(E/H) represents the deviation

between (u„ + u„) and 2u„, expressed in dB. In the near field of an antenna (or reflecting

Cj n h surface), it is well known that this error can be large, e.g., in the near field of an antenna

— 2 — ? one can find |h| /|e| ^ ". Such near-field problems are well known, and we shall not belabor

them. The question we address is the size of A(E/H) when there are a number of plane waves

present, i.e., multiple sources and/or reflections. The results for these situations are less

obvious and less well known.

The energy-density errors have been studied for both peak and average energy-density

measurements, for numbers of plane-wave sources ranging from two to twenty. For more than two

sources, it is impractical to use anything other than a statistical approach, since the error

depends on so many variables such as the relative strengths, phases, polarizations, directions

of arrival, and frequencies. Accordingly, for each number of sources we have generated 100 or 1000 random configurations (directions, magnitudes, etc.) and computed the mean, standard

deviation, and the extreme values of A , (E/H) and A (E/H).

pk avg

For the average energy density, there is no error if no two sources have the same

frequency. The errors are largest when all sources have the same frequency, as with multiple

reflections for example. Results for all sources having equal frequency are shown in Fig.

9. a. 5. For each number of sources, 1000 random configurations were generated, and the mean

and standard deviation of A (E/H) were computed for that ensemble. The typical errors were

from -2 to +2 dB, but the extremes were -10 to -12 dB and +3 dB. This was true whether the

sources were distributed in two or three dimensions.

131
For the error in the peak energy density, we used ensembles of 100 random configurations.

(The computation requires considerably more time than that for A .) Unlike A , A , (E/H)

avg avg' pk

is not zero when all the frequencies are different. Figure 9. a. 6 contains results for

A , (E/H). The mean of A , (E/H) tends to be around -0.3 dB with a standard deviation of 0.7 dB pk pk or less. If all frequencies are constrained to be equal, the standard deviation increases to

around 2 dB.

9. a. 2 Characterization of Complex Environments

A. General ities We now turn to the problem of how to characterize an EM environment.

In truth we are a long way from being able to do so comprehensively. The questions of what

are the quantities or characteristics of interest will depend on what must function in the

given environment. The important quantity could be the average total power density, the

maximum electric or magnetic field, the fraction of the time the field level exceeds certain

threshold value, etc. For some very important cases such as the human body, the quantity to

which it is most sensitive is not even known yet. A complete characterization of the EM

environment would amount to knowledge of the field at every point in space, which would

require measurements throughout the volume at a spacing of no more than one half wavelength

(for the maximum frequency present). That is usually impractical and almost never convenient.

A direct solution of Maxwell's equations is not feasible either, particularly since one typically does not know the source.

Given this situation, most current work on the problem attempts to determine some

potentially useful characteristics of the environment from a manageable number of measurements

[M]. There is as yet no really "good" (systematic, efficient) method which is fully

developed; but the field is still quite new, and progress is being made. We will discuss in a little detail a statistical approach to complex environments, and will briefly mention two

other lines of attack.

B. Statisti cal Approach Statistical methods (see for example [5]-[7]) are the only

methodical approach to be used in evaluating complex environments. In the absence of

information about radiating sources or propagation paths, one treats the sources or paths as random variables of position or time. (A good pedagogical treatment is contained in [5].) If

a large number of random sources contributes to the field at a given point and if the standard

deviation of any one source is not too large, then one can apply the central limit theorem. It basically states that the sum of an infinite number of random variables has a gaussian

probability distribution. Thus, if one component of the electric field is given by

E = X + jY = Z A. e-^*i , (9. a. 7)

where A. and $ . are random variables, then the probability distribution functions for the real

and imaginary parts of E take the form

p(X) = :^ exp[-(X - X)^/2 0^]

(2it)

(9. a. 8)

p(Y) = ~ exp[-(Y - Y)^/2 o^]

(2tt) Oy

where X, Y, o , o^ are determined by the distributions of the A. and ^.. In practice, these

distributions are unknown. Provided the phases ((>. are uniformly distributed, (9. a. 8) leads to

a Rayleigh distribution for the magnitude of E ,

P(|Ej) = ^exp[-|EjVa]

p(|E^| > E^) = exp(-E^ /a) , (9. a. 9)

„rms ,' E = /a

x 132
2 This is a remarkably simple result; it says that we need measure only one parameter (a = <E >)

in order to determine the complete probability distribution function. Such a result is

obviously too simplistic. Nevertheless, it does serve as a very useful first approximation,

and in some cases is even a good representation of the full answer.

There are numerous relevant examples in the literature, and two will be presented here.

Figure 9. a. 7 shows the results of measurements made at a large number of positions within a metal building which is irradiated from outside [8]. As is customary for such measurements,

the data are plotted on "Rayleigh paper" { - 0.5 log[- Z,n p(E > E )] vs. |E | in dB } . The

straight line representing the cumulative amplitude distribution obtained from a Rayleigh

distribution, (9. a. 9), clearly fits the data very well over the full range of measurements. A second example is shown in Fig. 9. a. 8, which is the result of measurements of the magnetic

field at one position (in a mine) at many different times [9]. (Note that the axes are interchanged relative to Fig. 9. a. 7.) Again, most of the data are represented very well by a Rayleigh distribution, although this time there is a clear departure at the low-probability

end (< ^%). Numerous other studies have been made of urban and suburban radio noise,

atmospheric noise, noise from automobiles, high-speed trains, power lines, etc. [5]-[7], [10]-

[16], all of which confirm the validity of the Rayleigh distribution for the bulk of the data.

This suggests that, in order to characterize a complex EM environment, one needs to measure E and the rest follows from (9. a. 9). As we indicated above, however, life is not

rms so simple. One problem is that, in all applications referenced, one determines E from the rms full distribution — no work is saved. This problem is probably surmountable. More serious

is the low-probability, high-amplitude portion of the curve, where the distribution deviates

from Rayleigh behavior. It is precisely this high-amplitude region in which our interest is

most likely to lie. Unfortunately, the characteristics of this troublesome region vary from

case to case, as can be seen from Figs. 9. a. 7 and 9. a. 8, for example. In fact this "lowprobability" region can extend up to 305^ to 50%; and its shape and maximum height vary as well. If the statistical approach is to be useful in EM environment characterization, the

deviations from Rayleigh behavior must be understood.

The origin of the unwelcome bump in Fig. 9. a. 8 is easy to understand qualitatively. If one had only the Gaussian background (9. a. 9), one would just get the Rayleigh amplitude

probability distribution represented by the solid line in Fig. 9. a. 9. If instead there were

just one strong source which happened to radiate 2% of the time always at a level 30 dB above some reference amplitude, it would appear as the dashed line in the figure. If the strong

source, instead of always having an amplitude of 30 dB when it was on, had a random amplitude

with the distribution peaked around 30 dB, then the corner indicated in the dashed line in

Fig. 9. a. 9 would be rounded, and the vertical and horizontal lines would be less steep and

less flat respectively. It is then easy to see how the combined effect of strong source plus

Gaussian noise could produce a distribution like that of Fig. 9. a. 8. The fact that the lowprobability bump is due to one or more strong sources which radiate some fraction of the time

(or influence some fraction of the positions) has led to its being called the impulsive

component of the noise.

A more quantitative presentation can be made by writing the field at some point as the sum of a dominant contribution, E , from the strong source plus contributions from all other

random sources, E , [5]. The probability density function for the dominant term, p (E ), is

not known — it could be a constant, normal, log-normal, whatever. From the central limit

theorem, the sum of the other contributions will have a Gaussian distribution for real and

imaginary parts, or a Rayleigh distribution for the amplitude,

^h 2 p^(E^) = —- exp(-E^ /a)

L I a L

The probability density function for the net component of the field is then

E = E , + E ^ x x d X I

2E

p(E ) = —- f exp[-(E^ + z^)/a] I (2E z/a) p.(z) dz , X a X o X d

133

(9. a. 10)
where I is the modified Bessel function. The limiting behavior for large or small E is then

p(E) = (2E/a) exp(-E^/a) , E^ « a,

(9. a. 11)

= P^(E) , E >> a,

where by E we mean \e \. Thus, the small-E (high probability) behavior is given by the

Rayleigh distribution, but the large-E impulsive component is controlled by the probability

distribution function of the dominant source(s). Since this impulsive component varies from

case to case, hope for a simple, universal statistical description fades.

More-complicated treatments have been developed, most notably by Middleton C17]-[19].

His interest is primarily in communications systems and in the effect of noise on the

receiver. He therefore distinguishes between noise which has a bandwidth smaller than that of

the receiver (Class A) and noise whose bandwidth is greater than that of the receiver (Class

B). For the case of multiple sources of impulsive noise (assumed to have a Poisson

distribution) plus a gaussian background noise, general forms for the resulting probability

distribution functions have been obtained in terms of a few parameters -- three for Class A and seven for Class B [17]. These general forms reproduced various measured probability

distributions very well, as can be seen in Fig. 9. a. 10. Unfortunately, there are rather severe restrictions on the general applicability of the formulae. The paper by Berry [18]

gives a clear account of both the derivation and the limitations of Middleton' s "canonical"

forms. The major unforeseen limitation is that the distance from observation point to the

first impulsive-noise source must be much greater than the path difference between the

observation point and any two sources --essentially, they must all be at about the same

distance. In addition, there is the requirement of Poisson statistics for the sources of

impulsive noise.

By including correction terms (and therefore more parameters) these restrictions can be

eased [19], but this approach is still not the general EMC tool we are seeking. The problem

is that there is virtually no predictive power. In all applications to date, the distribution

has been measured and the parameters determined from it rather than the reverse. What one has

then is a convenient representation of the noise-measurement results. This is very useful in

the design of communications systems (which was its purpose, after all), but it does not

enable one to characterize an EM environment by making just a few measurements. The same comment applies to other statistical quantities which have been employed, such as pulse-width

distributions, average crossing rates, Allan variance analyses, etc.: it is true that they

characterize the environment, but they require nearly complete measurement of the environment

to be determined.

C. Other Possible Approaches Although they are not yet at the production stage, other

approaches to complex environments are being investigated and developed. The goal of these

approaches is to extract some useful information about the environment from a relatively small

number of measurements.

An approach which holds some promise for large, open environments is directional scanning

[22]. The idea is to use a directional probe to measure the field in a number of directions,

distributed over the full Mir solid angle. These measurements can be used to determine the

coefficients in a (discretized) plane-wave expansion. In principle, once the coefficients

have been determined, the plane-wave expansion determines the field throughout the volume

scanned. In practice, the accuracy is limited by the fact that each measurement averages over some finite solid angle. Simulations indicate that the field reconstruction is rather poor,

but that useful upper bounds on the field can be obtained, and the rms field (where the

average is over the volume) can be determined quite well. Work still to be done includes

scanning in the presence of the earth or above a ground plane rather than in free space,

inclusion and assessment of measurement errors, various technical refinements, and eventually

practical tests.

Another method is based on a finite-element form of the EM action functional [4], [23].

The action is just an integral over position and time of some function which involves the

scaler and vector potentials. It has the property that, if the potentials at each point are allowed to vary, then a stationary point of the action with respect to this variation yields

potentials which lead to EM fields satisfying Maxwell's equations. In the approach being

134
developed, one first discretizes the volume of interest (for the sake of the computer) and

enforces boundary conditions and measurement results by fixing the potentials to be equal to the appropriate values at those points. The potentials are then varied at all other points

until a stationary point of action is found. The values of the fields which yield the

stationary point are solutions of Maxwell's equations which are consistent with the boundary

conditions and which have their measured values at measurement points. In practice, all this

is more easily said than done. Numerous technical difficulties arise; but none seems

insurmountable, and the work is progressing. This technique would complement the directional

scanning technique since the finite-element method is best suited to relatively small

(dimensions of order a few wavelengths or less), closed environments.

There are also other possible approaches which warrant some attention. It may be

possible to measure directly the parameters in Middleton's formula without measuring the full

distribution. In some situations, if the sources are local and identifiable, it may be

practical to characterize the radiating sources using just a few measurements, much as can be

done in EMC emissions testing [2M].

When it comes to characterizing a complex EM environment, there is no technique which is

systematic and efficient. Some environments have been characterized quite well statistically,

and these characterizations are useful for communications systems in particular. These

statistical characterizations, however, require extensive measurement effort and have

virtually no predictive power. The work being developed here will benefit significantly from

a better understanding of what are the EM characteristics of interest.

REFERENCES

[1] Randa, J; Kanda, M. Multiple-source, multiple-frequency error of an electric-field

meter. IEEE Trans. Antennas and Propag. AP-33(1): 2-9; 1985 January.

[2] Bassen, H. I.; Smith, G. S. Electric field probes - a review. IEEE Trans. Antennas

and Propag. AP-31(5): 710-718; 1983 September.

[3] Larsen, E. B; Ries, F. X. Design and calibration of the NBS isotropic electric-field

monitor (EFM-5), 0.2 to 1000 MHz. Nat. Bur Stand. (U.S.) Tech. Note 1033; 1981 March.

[M] Kanda, M; Randa, J.; Nahman, N. S. Possible estimation methodologies for

electromagnetic field distributions in complex environments. Nat. Bur. Stand. (U.S.)

Tech. Note IO8I; March 1985. '

[5] Beckman, P. Probability in Communication Engineering. New York, NY: Harcourt, Brace,

and World; 1967.

Skomal , E. Man-Made Radio Noise. New York, NY: Van Nostrand Reinhold; 1978.

Lee, W. C. Y. Mobile Communications Engineering. New York, NY: McGraw-Hill; 1982.

Hoffman, H. H. ; Cox, D. C. Attenuation of 900 MHz radio waves propagating into a metal

building. IEEE Trans. Antennas and Propag. hP-30{H): 808-811; 1982 July.

Kanda, M. Time and amplitude statistics for electromagnetic noise in mines. IEEE

Trans. Electromag. Compat . EMC-17(3): 122-129; 1975 August.

Haskard, M; Miller, M. ; Johnson, A.; Marconi, P. Radio noise measurements in an urban

environment. J. Elec. Electron. Eng., Australia, 2, no. 2: 9^-102; 1982 June.

Struzak, R.; Rymarowicz, Z; Moron, W. Urban MF radio noise survey in Poland. Proc.

5th Symp. and Tech. Exhibition on EMC; 1984 March; Zurich, Switzerland. 1-6.

Parsons, J.; Sheikh, A. Statistical characterization of VHF man-made radio noise.

Radio Elec. Eng., 53(3): 99-106; 1983 March.

Nakai , T. ; Kawasaki, Z. I. Automotive noise from a motorway: I, measurements. IEEE

Trans. Electromag. Compat. EMC-26(4): 169-174; 1984 November.

Shepherd, R. Measurements of amplitude probability distributions and power of

automobile ignition noise at high frequency. IEEE Trans. Vehicular Tech. VT-23(3): 72-

83; 1974 August.

135

[6]

[7]

[8]

[9]

[10]

[11]

[12]

[13]

[14]
[15] Nakai , T. ; Kawasaki, Z. I. APD's and CRD's for noise from bullet trains in Japan.

IEEE Int. Symp. on EMC; 1984 October; Tokyo, Japan. 250-255.

[16] Sheikh, A.; Parsons, J. Statistics of electromagnetic noise due to high-voltage power

lines. IEEE Trans. Electromag. Compat . EMC-23(4): 412-^419; 1981 November.

[17] Middleton, D. Statistical-physical models of electromagnetic interference. IEEE

Trans. Electromag. Compat. EMC- 19(3): 106-127; 1977 August.

[18] Berry, L. Understanding Middleton's canonical formula for Class-A noise. IEEE Trans.

Electromag. EMC-23(4): 337-3^^; 1981 November.

[19] Middleton, D. Canonical and quas i

canoni cal probability models of Class-A

interference. IEEE Trans. Electromag. Compat. EMC-25(2): 76-106; 1983 May.

[20] Adams, J. W. ; Bensema, W. D.; Kanda, M. Electromagnetic noise: Grace Mine. Nat. Bur.

Stand. (U.S.) Int. Rept. 7^-388; 1974.

[21] Bolton, E. C. Simulating LF atmospheric radio noise at 60, 76, 200 kHz. Office of

Telecommunications, OT-TM-97; Boulder, CO. 1972.

[22] Randa, J.; Kanda, M. Directional scanning of complex electromagnetic environments.

IEEE Trans. Antennas and Propag. AP-33, no. 12: 1413-1416; December 1985; Also, A directional scanning technique for characterization of complex electromagnetic

environments. Proc. IEEE Int. Symp. Antennas and Propag.; 1985 June; Vancouver,

Canada

.

[23] Randa, J.; Kanda, M. A finite-element action approach to the characterization of

complex electromagnetic environments. Proc. Int. Symp. Antennas and EM Theory; 1985

August; Beijing, China.

[24] Ma, M. T. ; Koepke, G. H. A method to quantify the radiation characteristics of an unknown interference source. Nat. Bur. Stand. (U.S.) Tech. Note 1059; October 1982;

Also A new method for determining the emission characteristics of an unknown

interference source. Proc. IEEE Int. Symp. EMC; 1982 September; Santa Clara, CA. 151-

156.

136
Figure 9.a.l Equivalent circuit for a single antenna plus

diode, with filter line also indicated.

400 1000

Er*(V/m)

Figure 9. a. 2 Error in average electric field strength for

E^(t) = E^rms

(cos cot + cos 2a)t>. Error in

peak strength is given by A , - A - 3 dB, pk avg

137
400 1000

E^^V/m)

Figure 9.a.3 Error in average electric field strength for a periodic square wave, + ETM®. In this case. pk A + 3 dB, avg

(D

10

8-

6-

4-

2-

-1

1 1 1 II 1 ^

8/p — = ao5

— /

- / ^^

8/P - 0.1

'^

m /^ —

— / / — ^

8/p - 0.2 —

- \-^.y^ —

1 II II 1 10 40 102 400 lO'

E['~(V/m)

Figure 9. a.

4

Error in average electric field strength for a periodic square pulse with various ratios of wid th

(5) to period (p) . Peak value is EP^^ = ET^^/pT^

.

The error in the peak strength is given by

A , - A = - 10 dB, -7 dB, -A dB for 6/p = 0.05, pk avg ' ' ^ ' 0.1, and O.Z respectively.

138
1 -

-X —»

X X X

ffi B

Z

^ -1

-2

•3

-4

(i M

2

J AVQ ± c

" Max

' '

I

' I I '

I

' t'

t

I 6 8 10 12 14 16 18 20

••ourca

Figure 9. a. 5 Mean, standard deviation, and maximum values

of the error A^„„(E/H) , as functions of the avg

number of sources. For each N , 1000 source random configurations were used.

u

2-

1-

a

1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 Avg ±<r_

xMax

* * X » * » X

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f

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< .2

3

2 4 6 8 10 12 14 16 18 20

Nftourca

Ficure 9. a. 6 Mean, standard deviation, and maximum and minimum

values of A (E/H) , using 100 random configurations

for each Nsource

139
OlSTRlSUTtON

TOTAL SUILOINC

^ ncremieo to tNCiocNT ncLO ^ LCVCL AT CACH HCiCHT

• ncrcKiiCD TO HCICHT -AveiiAceo

IMCIOCNT nCLO

«90 «45 -40 -35 -30 -25 -20 -15

RCLATlve SIGNAL LCVCL i4B)

Figure 9. a. 7 Overall signal-level distribution, from [8].

140
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EM noise in a mine [9].

141
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Figure 9. a.

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142
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144
9.b OUT-OF-BAND EMC PROBLEMS

Interfering signals at out-of-band frequencies can be either intentional, as in jamming

with high-power microwave beams, or unintentional, as in typical EMC problems. In either

case, it is important that the EMC analyst be able to understand and to predict out-of-band

coupling phenomena. The importance of the out-of-band response of antennas [1],[2] and antenna

feed systems [3] in EMC problems has been recognized for some time.

Our initial research at NBS in this area has concentrated on the out-of-band response of

reflector antennas [4]. Reflector antennas are of particular interest because they are used

so frequently at microwave frequencies and because they have a strong response to above-band

frequencies. The characterization of the above-band response of reflector antennas is

complicated because of the presence of higher-order propagating modes in typical waveguide

feeds. This problem is avoided in the receiving-mode analysis presented in Section 9.b.1,

where the total received power carried by all the propagating modes is computed. The power

coupled from the waveguide to the detector depends on the specific feed system, and, in

Section 9.b.2, we analyze the out-of-band response of a typical coax- to- waveguide adapter. We

also combine the theoretical results for reflector antennas and coax-to-waveguide adapters for

comparison with measured data at out-of-band frequencies [2]. Section 9.b.3 contains

conclusions regarding reflector antennas and a brief description of our present work on phased-array antennas.

9.b.1 Reflector Antennas

Reflector antennas typically use waveguide feeds, and frequencies well below the in- band

frequency are not important because they are cut off by typical waveguide feeds.

Consequently, "out-of-band" will refer only to above-band frequencies. We choose to analyze

reflector antennas in the receiving mode using a two-step method. In the first step, we use the physical optics approximation to compute the fields in the focal region of a parabolic

reflector. In the second step, we integrate the Poynting vector over the aperture of the feed

horn to determine the total received power. The mathematical details are given in [^], and

here we will just describe the general procedure.

A perfectly conducting, paraboloidal reflector of diameter D and focal length f is shown

in Fig. 9.b.l. The origin of a rectangular coordinate system (x, y, z) is located at the

focus of the paraboloid. A plane wave is incident at an angle 9 to the z-axis in the xz

plane. The physical optics surface current J on the reflector is given by

J = 2 n X H. , (9.b.1)

Where n is the unit normal to the reflector and H. is the incident magnetic field. An infinitesimal surface current patch of area dS at a point P, on the reflector produces

electric and magnetic fields [5], [6] at the point P :

-JkTiQ g-JkR

(9.b.2)

-jkR

dH = 4^ (J X u^) ^-5— dS — 4ii — —

R

Where t\^ is the intrinsic impedance of free space. As indicated in Fig. 9.b.l, R is the

distance between P, and P_, and u^, is the unit vector directed from P to P . To compute the

fields in the vicinity of the focal region, it is necessary to integrate (9.b.2) over the

surface of the reflector. In general, numerical integration is required; but simple,

approximate expressions for the electric and magnetic fields, E and H, and the Poynting

* vector, S = Re(E x H )/2, have been obtained [^],[6]. Here Re indicates real part, and * indicates complex conjugate.

A waveguide feed and a feed horn whose aperture is centered at the focal point of the

paraboloid are shown in Fig. 9.b.2. Normally, the feed-horn aperture dimensions are on the

145
order of a wavelength at the in-band frequency. Consequently, the aperture dimensions can be

assumed to be electrically large at above-band frequencies, and we can make the Kirchhoff

approximation for the aperture field [4]. The received power is then given by the integral of

the Poynting vector over the aperture, A,

- ^A \

dA (9.b.3)

where S is the z-component of the Poynting vector, and the geometry is shown in Fig. 9.b.2.

Since we assume that no power is dissipated in the walls of the horn and waveguide, the power

propagating down the waveguide is also given by P in (9.b.3). Because the waveguide will

normally be multimoded at above-band frequencies, P is the total received power in all the

propagating waveguide modes. For general feed-horn apertures, the integration in (9.b.3) must

be done numerically; but for circular apertures, a simple approximation has been obtained for

P [^].

r

We can define a generalized effective aperture A as the total received power divided by

the incident power density,

1 A = P / ( _ e r 2 ^0 l"il ) (9.b.4)

It is often convenient to normalize A to the physical area of the reflector antenna, A = e ^ ' p

irD^/il For the case of on-axis incidence (6 = 0) where A is maximum, A /A can be

s e e p approximated by ['4],[5]

A /A = 1 - J^(ke p ) - J^(ke p ) e p m^m 1 m^m (9.b.5)

where k is the free-space wavenumber (2ir/X), 6 is the angle defined in Fig. 9.b.l, p is the m m radius of the feed horn, and J and J are zero-order and frist-order Bessel functions The

ratio A /A is sometimes called aperture efficiency and is less than unity. For sufficiently

e p

p J high frequencies (ke p >> 1), A /A approaches unity. mm e p

For the off-axis case ( 0) , an approximation similar to (9.b.5) has been obtained

[4]:

A /A =1 e p

1/2 [2 - j^(ke„p^) - jf(ke„p^) - jf^ke^p ) - J?(ke^p )] m + 1 m + m

- 1 m-

f sine < p s m

4 /^ [Jn^ke^p ) + J?(ke^p ) - J^(ke^p^) - jf(ke„)]

c m

-

1 m

- m + 1 m

f sine > p s m

(9.b.6)

where

f sine ± p s ^m' and sin (p /f sine ) ^m s

This approximation yields the receiving pattern and the beamwidth. Measured receiving

patterns have been published for a 3 GHz reflector antenna at frequencies up to 10 GHz [2]. A comparison of theory [4] and experiment [2] is shown in Fig. 9.b.3 for a frequency of 6 GHz.

The agreement in Fig. 9.b.3 is typical of that obtained at other frequencies. The theory

which represents total received power in all the modes tends to provide an upper-bound

envelope for the measured power.

146
A number of extensions to the basic reflector-antenna analysis have been made [4]. The

transient response has been studied by inversely transforming the frequency-domain expressions

to the time domain. The effect of reflector roughness has been examined using the classical

result from Ruze [7]. Offset parabolas and dual reflectors have also been treated by using the

equivalent diameter and equivalent focal length concept of Hannan [8].

9.b.2 Coax- to Waveguide Adapter

In many out-of-band EMC problems, the response of the feed system is at least as important as the antenna response. In attempting to interpret the out-of-band measurements of

Cown et al. [2] on a 3-GHz reflector antenna, we found that we needed to know the out-of-band

response of the coax-to-waveguide adapter which was used in that feed system. Such adapters

are also used with many other antennas which employ waveguide feeds, and its frequency

response illustrates the complexity of the out-of-band problem [^i].

The probe-type adapter which we consider is shown in Fig. 9.b.4. Collin [9] has analyzed

this structure using a variational technique, but his computational technique is valid only

for in-band frequencies where the waveguide supports only a single propagating mode. We have

extended his analysis to allow for higher-order propagating modes which occur at above-band

frequencies [^1]. Here we will show some typical results, but will skip the mathematical

details given in [4].

If the coaxial cable is sufficiently small, then it will support only the dominant TEM

mode even though the waveguide supports higher-order propagating modes. The main task is to

compute the input impedance Z. of the probe as seen from the junction with the coaxial cable

(y•'in = 0). Z. can be written as a mode sum,'

00 00 Z. = R. + j X. = I I Z (9.b.7) in in ^ m , -, r- r^ TM^

n=1 ,3,5 m =

where Z = R + J X , nm nm nm

R is the resistance associated with the nm waveguide mode, and X is the reactance

nm nm

y associated with the nm waveguide ° mode. The probe ^ excites only TM modes that are transverse

nm

Y magnetic ^ to y.' The fields of the TM mode have n half cycles in the x direction and m half

nm cycles in the y direction. The terms for n even do not contribute to Z. because the probe is

located at the center of the waveguide (x = a/2). Only the propagating modes contribute to

the real part *^ of Z. , and Collin [9] treated the case where only the TM;„ mode was

1 n ^10

propagating. The double sum in (9.b.7) can be transformed to improve the numerical,

convergence [4].

In considering the power transmission from the coaxial cable to the waveguide, we assume

that the coaxial cable has a real characteristic impedance R . Then the voltage reflection

coefficient, r , is given by

r = (Z. - R )/(Z. + R ) (9.b.8)

v in c m c

The power reflection coefficient is then given by I r I , and the transmission coefficient T

V L- for the total power supplied to the waveguide is

o 4R. R T^ = 1 - \t/ = '^~ (9.b.9)

^ ^

(R. + R )'^ + X^

in c in

147
We can also define the transmission coefficient for the nm mode T as the ratio of the power

nm transmitted to the nm mode to the incident power in the coaxial cable, and T is given by [i<]

4R R T = "^^^ ^ . (9.b.10)

"•" (R. - R )2 . X^

in c in

The total transmission coefficient T, can also be written as the sum of the modal transmission

coefficients

T, = I Z T . (9.b.11)

t nm

n m

The sum in (9.b.11) is always finite because only a finite number of waveguide modes propagate

and carry power.

In the receiving antenna problem, we are interested in the reciprocal problem where a propagating waveguide mode transmits power to the coaxial cable. By reciprocity, we can show

that the power transmission coefficient T as given by (9.b.lO) also applies to this case. nm

y In Fig. 9.b.5, we show the transmission coefficients for the dominant mode (TM:: ) and the

y first higher-order propagating mode (TM': ). The adapter parameters were chosen to match those

used in the reflector antenna that was studied experimentally by Cown et al . [2], [10]: a = 7.112 cm, b = 3.302 cm, d = 1.9 cm, i = 2.^ cm, t = 7.0 mm, and R = 50 n. The in- band

frequency is 3 GHz, and T is nearly unity from about 2.5 to 5.0 GHz. At 5.0 GHz, the first

higher-order mode appears, and other higher-order modes appear above 6 GHz.

In an earlier comparison between theory and experiment for the on-axis gain of the

antenna, the experimental values were well below the theoretical results. The main problem

with the comparison was that the theory included only the antenna response, while the

experimental included the antenna plus the adapter response. If we take the total

transmission coefficient T as given by (9.b.9) or (9.b.11) and multiply it by the effective

aperture of the antenna, then a corrected frequency response is obtained as shown in Fig.

9.b.6. To convert from effective aperture to gain, we divide A by the effective aperture for

2 an isotropic antenna (X /4it). No precise correction can be made because we do not know the

modal content incident on the adapter from the waveguide, but the adapter correction does

improve the agreement significantly, particularly below 7.5 GHz. Above 7.5 GHz, the adapter

theory is probably not valid because the adapter probe no longer behaves like a thin, linear

antenna.

Other adapters, such as those with loops rather stubs, could be studied theoretically or experimentally at out-of-band frequencies. Many other antenna types and feed systems are also

worthy of out-of-band analysis or measurement. Recent unpublished experimental results with a variety of antenna types show fairly large mismatch losses at out-of-band frequencies [11].

Our present out-of-band work at NBS involves the analysis and measurement of large

phased-array antennas. These antennas have rather complicated patterns [12] because of the

presence of grating lobes at above-band frequencies.

REFERENCES

[1] Lind, W. R. Out of band performance of antennas. Moore School of Electrical

Engineering Report No. 65-I8; 1965.

[2] Cown, B. J.; Ryan, C. E. ; Weaver, E. E. Antenna pattern measurements to characterize

the out-of-band behavior of reflector antennas. Georgia Institute of Technology,

Engineering Experiment Station, Report ECAC-TR-83-OO3 ; 1983.

[3] Levinson, D. S. Transmission-line measurement technique. Rome Air Development Center

Technical Report No. RADC-TR-65-I 87 ; 1965.

148
[i<] Hill, D. A. Out-of-band response of reflector antennas. Nat. Bur. Stand. (U.S.) Int.

Rept. 85-3021 ; 1985.

[5] Minnett, H. C.; Thomas, B. M. Fields in the image space of symmetrical focusing

reflectors. Proc. lEE, vol. 115: U19-1430; 1968.

[6] Valentino, A. R.; Toulios, P. P. Fields in the focal region of offset parabolic

antennas. IEEE Trans. Antenna Propag. AP-2i1(6): 859-865; 1976 November.

[7] Ruze , J. Antenna tolerance theory - a review, Proc. IEEE, vol. 5^(4): 633-640; 1966

April.

[8] Hannan, P. W. Microwave antennas derived from the Cassegrain telescope. IEEE Trans.

Antennas and Propag. AP-9(2): UO-153; 1961 March.

[9] Collin, R. E. Field Theory of Guided Waves. New York, NY: McGraw-Hill, I960; sec.

7.1.

[10] Cown, B. J.; Ryan, C. E. Near-field theory and techniques for wideband radiating

systems at in-band and out-of-band frequencies. Georgia Institute of Technology,

Engineering Experiment Station, Contract No. DAAG29-78-C-0029; 1982.

[11] Hill, D. A.; Francis, M. H. Out-of-band response of antenna arrays. Nat. Bur. Stand.

(U.S.) Int. Rept. 86-30^17; 1986 June.

[12] Cown, B. J; Cain, F. L. ; Duffy, E. G. Statistical prediction model for EMC analysis of

out-of-band phased array antennas. IEEE Trans. Electromag. Compat. EMC-l8(i4): 163-170;

1 976 November

149
P2(X2.Y2.0)

Figure 9.b.l Geometry for a plane wave incident on a symmetrical paraboloidal reflector.

Feed Horn

/

Waveguide

r Pr

Figure 9.b.2 Geometry for the feed horn and waveguide,

150
-10-

-20

-o

N-^ k.$

1

-30

0.

>

to -40

0)

QC

•50-

-60

-40

e<

Figure 9.b.3 Comparison of theory [A] and experiment [2] for

the above-band frequency of 6 GHz.

Antenna parameters are: D = 1.22 m,

f/D = 0.32, and pj^ = 3.39 cm.

151
*y

/

T

d

1

Figure 9.b.4 Probe type of coax-to-waveguide adapter,

152
Figure 9.b.5 Transmission coefficients for an S-band coax-towaveguide adapter. The cutoff frequencies for

the dominant (TM^q) and first higher order (TM^^)

modes are indicated by arrows.

153
50

40

20

10-

••o*" OO

With Adapter Correction

Cown et al. • • • •

5 6 7

Frequency (GHz)

8 10

Figure 9.b.6 Comparison of theory (with adaoter correction) and

experiment [2] for antenna gains as a function of

frequency.

154
9.C CONDUCTED EMC PROBLEMS

As part of the overall EMC goal in an undesired-source environment, one often encounters

what are mainly conducted EMI (CEMI) problems. Figure 9.C.1 depicts the basic interference

mechanisms between an emitter and a susceptor [1]. The undesired coupling due to radiation is

indicated by Paths 1, 2, 3, and 4. Perhaps the most common case of CEMI is the direct

conduction of interference via the power line shown as Path 5. However, as indicated in Fig.

9.C.2, other less obvious forms of CEMI occur due to conductive coupling via common ground

lines and capacitive and inductive transfers. These paths generally provide well-defined

transfer functions, such that the coupled interference can be determined in terms of stray

capacitance (C), mutual and self inductances (M and L) , and coupling resistance (R). However,

leakage paths, pick-up, and reradiation affect the total amount of interference that would

otherwise be transmitted along a conducted path. The distinguishing feature of conducted

interference is that a conducted path is the primary means by which the source is connected to

the susceptor [2].

Thus, the CEMI problems can be approached by examining the three basic areas of emitters,

susceptors, and the coupling or transmission paths which inadvertently combine to cause interference. The following discussion provides a brief description of some CEMI problems

that are representative of the present attempts to achieve compatibility in the EM

environment

9.C.1 Characterizing CEMI Sources

Unwanted conductive emissions originate in ac power systems due primarily to the direct

transients caused by switching reactive loads, relay or breaker contact bounce and restrikes,

thyristor (solid-state) phase control, etc., and to induced transients caused by lightning

discharges, the coupling from cordless phones, computers, RF-energized lightbulbs, and

motorized equipment. One of the fundamental CEMI problems, therefore, is to characterize and

model (if possible) the nature of these transients on ac power systems.

An appreciable body of data exists which provides a basis for characterizing the

transient disturbances likely to be encountered on land-based ac power systems intended for

industrial and residential use [3]-[5]. Although the exact amplitude, waveshape, and duration

of these disturbances depend on the cause (lightning, switching transient, induced RF, etc.)

and location in the power system (generator, main transmission line, substation, local

distribution line, etc.). Fig. 9.C.3 shows a compilation of the frequency of occurence vs. peak amplitude of voltage transients that occur in 220-V and 120-V systems [1]. Obviously,

transients often can occur which are 10 times the nominal peak voltage and sometimes can be

nearly 100 times larger.

Of primary interest, of course, is the kind of overvoltage transients that can occur at the local distribution transformer, service entrance, and branch circuits of buildings and

homes. A considerable amount of study has gone into this problem in order to protect

transformers, switchgear, and consumer appliances from damage due to transient "surges" [M]-

[9]. From these studies has come a set of test waveforms whose amplitudes, shapes, and

durations represent (by consensus) the actual multitude of random, arbitrarily- shaped

(conductive) interference waveforms that can appear on an ac power line (see Fig. 9.C.4).

Figure 9.C.5 shows the IEEE Std. 587 (now designated ANSI/IEEE C62. 41-1980) consensus

waveforms and where they are applicable for testing purposes relative to the ac power feeder

line [10]. Similarly, Subcommittee 28A of the International Electrotechnical Commission (lEC)

has prepared a report in which "Installation Categories" are defined [8].

Nevertheless, problems do remain in that there is still a lack of enough definitive data

on the duration, waveform, and source impedance of transient disturbances in land-based ac power circuits. Consequently, attempts to accurately model ac power systems for CEMI

analyses, or for component design purposes, are difficult and cumbersome. Bull [11] reports

that the impedance of an ac power system seen from the outlets exhibits the characteristics of

a 50-fi resistor with 50 yH in parallel. However, attempts to combine the IEEE Std. 587 6-kV

open-circuit voltage with the assumption of a 50-J2/50-viH impedance resulted in low energy

deposition capability in a transient voltage suppression device, contrary to the field

experience of these devices [7]. Also, there are some significant differences between the

concepts of the IEEE Std. 587 and the lEC 664 report. At present, source impedances (and,

thus, current levels) have not been defined in the 664 document.

The problem of characterizing transient disturbances of shipboard power systems has

become critical as electronic systems more susceptible to transient interference are placed on

155
the ship powerline buses. The following quote is taken from a recent paper on this subject

[12]:

"Digital systems are particularly susceptible to this wideband

interference. It is suspected that many computer programs are reloaded

too often because their electronics and/or their signal cables have been

exposed to transients, which are either conducted on, or radiated from,

powerline buses. Presently, a meager data base exists to describe the

magnitude and other characteristics of these powerline transients."

In contrast to land-based ac power systems, there has not been very much information

gathered concerning the characteristics of shipboard ac power systems, probably because of the

varying configurations and performance requirements for different classes of ships. With the

advent of transistors and solid-state electronics appearing in shipboard equipment during the

1960s, staff at the David Taylor Naval Ship Research and Development Center (DTNSRDC) in

Annapolis, MD began to concern themselves with transients on shipboard power systems. Some

definitive work in establishing a baseline of statistical data was carried out in the late

1960s to gather information on the magnitude of transient disturbances and their frequency of occurrence [13]. Figure 9.C.6 shows a seven-day histogram of the number of transients vs. peak amplitude taken on ships having both 120- and J440-V systems. Note the similarity with

land-based frequency of occurrence transients given in Fig. 9.C.3. It appears that the

magnitude of shipboard ac power line disturbances and their occurrences correspond

approximately to what is considered a low exposure level for land-based systems [7]. This

seven-day histogram subsequently became incorporated as part of DoD-Std-1399 [14]. It is

interesting to note, however, that the frequency of occurrence histograms given in [12] (taken

in 1983) show peak amplitudes considerably lower than the data taken in 1960s (see Fig.

9.C.7).

The lack of a statistically significant data base to characterize the transients on shipboard power systems became problematic when in August, 1982 a program to develop

recommendations for voltage "spike" suppressor specifications for Navy ships was undertaken by

NBS [15]. The objective of the program was to provide a sound technical basis for shipboard

application of transient suppressors, based on analysis of existing field data gathered from

shipboard testing. Subsequent investigation of available reports and other data gathered as part of the overall Navy Electric Power Interface Compatibility (EPIC) program revealed that

the existing data were far from adequate to develop a meaningful specification.

The EPIC program was established to identify, if possible, the sources of incompatibility

between shipboard power systems and user equipment. Some of the main areas of study were

voltage interruptions, voltage drops, harmonic distortion, and voltage spikes. For the

voltage-spike area, the EPIC test procedures called for specific measurements of spike

amplitude, waveshape, frequency of occurrence, and broadband impedance. These quantities are needed in order to determine the peak current and total energy dissipated in a clamping type

of suppressor. Figure 9.c,8 shows two examples of the effect of clamping- type suppressors on transients superimposed on a steady-state-power voltage wave. The top example (a) has the

clamping level set correctly, whereas the bottom example (b) has the level set too low.

Selecting the steady-state voltage rating is obviously important so that the suppressor does

not have to absorb excessive amounts of energy in the absence of spikes 1,3, and 5. Figure

9.G.9 shows a circuit for calculating both the peak current and the energy dissipated in a clamping suppressor due to voltage spike, V , superimposed on a steady-state voltage, V . A total voltage, V + V , is thus imposed upon a circuit which is protected by a clamping

suppressor having a clamping voltage V , through a source impedance Z . Current i then

exists when V + V > V . During this time, the suppressor must absorb energy, E : s ss c s

^2

E = V / i dt , (9.C.1)

s c ' s

' ^1

where t,12to t^ is the time interval during which V + V > V , and V is assumed to be

s ss c c constant (ideal suppressor). Since it is difficult to measure i in practice, an alternate

method of determining E is to use a recording of the voltage spike together with the measured

156
source impedance as a function of frequency. Then, since i = (V + V - V )/Z , (9.C.1)

5 S o S OS

becomes

t^ V (V + V - V ) E^ = ;

2 c s ^ ss c_

^^ ^ ^^^^^^^

where V > V , and Z is assumed to be constant between times t, and t^.

c ss s 12

From the above discussion, it is apparent that data on spike voltage waveshape and

duration are important, as are source impedance data, in determining realistic suppressor

specifications for energy absorption and peak current ratings. Although a few records were obtained from several ships showing spike voltage waveforms, and a very limited amount of

impedance data was taken on two ships, the information available from EPIC program at that

time contributed minimally to the development of a possible voltage-spike-suppressor

specification. A broadband impedance measuring system (BIMS) was developed for the Navy by a private contractor which can measure the complex impedance at multiple points in a shipboard

power system without de-energizing the line or interfering with it. Some of the initial

results obtained with this device were reported by NBS [15].

9.C.2 Reducing CEMI Coupling by Grounding Computer Facilities

The principles of good grounding practice are generally well known and described in the

literature [2], [5]. However, for the installation and operation of many automatic data

processing (ADP) facilities, the problems of CEMI coupling have become significant towards

achieving adequate system performance. A special conference on the subject of "Power and

Grounding for a Computer Facility" was held at NBS, Gai thersburg , MD in May, 1984. The

Institute for Computer Science and Technology at NBS has recently issued a federal information

processing standard (FIPS) publication describing techniques of electrical power for ADP

installations which deals, among other topics, with problems of grounding [16].

Three fundamental grounding concepts are shown in Fig. 9.C.10. The floating or isolated

ground technique is, of course, the most effective means for preventing current in one circuit

from coupling into another circuit. Transformer and optical isolation methods are often used

for this purpose where the effects of stray winding and case capacitances are minimized by means of electrostatic shields. Single-point grounding is the method by which all groundreturn currents are routed to a single reference potential (typically, earth ground), thus

preventing any unwanted voltage drops between separate circuit grounds. Where large amounts

of line power are supplied to equipment, such as in an ADP facility, maintaining a good

single-point ground at the entry for utility, telecommunications, and ADP power is essential.

Multipoint grounding is possible where (ideally) a zero-potential, zero- impedance ground plane

is available. At frequencies where ground leads and cable lengths become comparable to signal

wavelengths, single-point grounding becomes impractical, and multipoint grounding is used to

minimize ground lead lengths. If possible, a large conductive body is used for the ground

plane to minimize ground loops and equalize "ground potential".

Because of the trend toward increased speed (higher signal bandwidths) and lower signal

levels in ADP equipment, computer facilities face stringent grounding requirements. For

avoiding CEMI problems, most ADP centers use the approach shown in Fig. 9.C.11 which

illustrates several concepts. First, as mentioned above, all electrical services required are located together where interconnecting conducting straps between service-entrance panels can equalize ground voltage differences (for frequencies up to 1-10 MHz). Second, to keep power,

neutral, and ground leads short (10 meters or less), a power center (typically, isolation

transformer) is used to service each cluster of ADP equipment. This unit provides another

degree of single-point and isolated grounding, minimizing any noise-voltage differences that

may exist between the service-entrance earth ground and the local room structure grounds.

Third, a raised floor support structure is often employed which is intended mainly as a multipoint ground (or signal reference grid) for equalizing potentials between ADP units from

either radiated or conducted emissions having frequencies above 1-10 MHz.

Quantifying the effectiveness of these grounding measures remains a problem.

Differential and common-mode currents generated by nonlinear loads in ADP equipment can propagate back through isolation transformers and into other victim equipment. These currents

collect in ground-plane structures and must be measured with special current probes. The

design of such probes and their placement for obtaining a profile of CEMI coupling effects is

the subject of present research and development efforts [1]. Noise transients couple into

157
computer systems via rectification processes in dc power supplies, by discharge from

statically charged human bodies, via interconnecting data cables, and through small potential

differences induced in multipoint grounds. Test- equipment noise simulators are now available

Which claim to be capable of determining noise voltage and current threshold levels in ADP

facilities and other field environments [17].

9.C.3 Measuring CEMI Emission and Susceptibility Levels

A. Emission Testing Because of FCC EMI regulations which must be complied with as of

October, 1983, conducted emission limits and the means for testing to meet these limits have

become the subjects of many trade-journal articles [l8]-[20]. Figure 9.C.12 illustrates the

basic conducted emission conditions (noise feedback into the ac power line) that the FCC

regulations are aimed at controlling. With the equipment under test powered on, together with

associated connecting cables, any conducted coupling (via stray reactances, etc.) can cause common- and differential-mode CEMI currents, I and I , to be generated on the power line

cord. Since the magnitude of these currents depends on the varying ac power line source

impedance, the FCC Part 15 Rules establish test procedures that make use of a line- impedance

stabilization network (LISN) which has an impedance characteristic as shown in Fig. 9. c. 13.

Various designs for LISN circuits can be used to achieve the desired characteristics. The

conventional 50-fi, 50-yH LISN characteristic is also shown in Fig. 9.c,13, However, a twosection LISN recommended by the VDE also prescribes an impedance characteristic down to 6 n at 10 kHz [19] (see Fig. 9.0.1^4). As indicated in Fig. 9.C.14, conducted emission testing is

accomplished by observing the common- and differential-mode peak voltages measured with a spectrum analyzer having a controllable bandwidth [19]. Alternatively, ANSI/IEEE C52.2 calls

for a radio-noise-meter measurement in place of spectrum analyzer. Evaluating the conducted

emissions from a piece of equipment would ideally require a virtually noise- free environment.

A shielded room is recommended (although the FCC does not use one [20]), together with heavy

filtering of the ac power lines, and test equipment with noise levels below the specification

limits. Because of pickup on the power-line cord (acting as an antenna) by emissions

(generally above 10 MHz) from the I/O cables, cable placement should be varied, keeping the

line cord rigid, to obtain the highest reading conditions.

Probably the biggest problem in conducted emission testing is keeping tabs on the various

FCC (USA), VDE (West Germany/European Common Market), CSA (Canada), SEV (Switzerland), and lEC

specifications. Also, in selecting the line filter and/ or line capacitors for attenuating

noise currents (I and I in Fig. 9.C.12), the allowable safety limits shown in the following

table require tradeoffs to be made [19].

Table 1: Capacitive Current Limits for Grounded Equipment

Regulation Limits

United States UL 478

UL 1283

Canada CSA 22.2 No. 1

Switzerland SEV 105^^-1 lEC 335-1

West Germany VDE 0804

5 mA,

120 V, 60 Hz.

0.5 to 3.5 mA,

120 V, 60 Hz.

5 mA,

120 V, 60 Hz.

0.75 mA,

250 V, 50 Hz.

3.5 mA,

250 V, 50 Hz.

158
For example, a capacitor that draws a 3.5 mA ground current from the line also provides 15 dB

more attenuation to noise emissions above 1 MHz than a capacitor drawing only 0.5 mA of the

ground current. Thus, meeting both safety restrictions and passing acceptable conducted

emission testing levels (which are subject to change) presents serious considerations. For

military applications, there are also a number of EMC standards dealing with component and

system performance. MIL-STD-M61 B, "Electromagnetic Emissions and Susceptibility Requirements

for Control of EMI", defines emission and susceptibility limits applicable under various test

conditions. Figure 9.C.15 shows the present and proposed conducted emission levels set by the

FCC and VDE for both consumer (Class B) and industrial (Class A) "digital" equipment. [FCC

Docket 20780 defines equipment in this category as "a device or system that generates timing

pulses at rates in excess of 10,000 pulses (cycles) per second and uses digital techniques."]

Class B devices comprise computing equipment — personal computers and peripherals, electronic

(TV) games, calculators, watches, etc. — marketed for home usage. Class A devices include

computing products intended for use in commercial, industrial, and business environments.

Presently exempt from FCC Part 15 Rules are automotive electronics, industrial control

systems, test equipment, medical equipment, and home appliances. FCC conducted emission

limits start at 0.^5 MHz and extend to 30 MHz, while the VDE regulates down to 0.15 MHz and is

proposing limits down to 10 kHz. The upper VDE 0871A specifications (for industrial devices)

do not allow the additional 9.5-dB limit above 1.6 MHz permitted by present FCC levels.

One more hitch to the conducted emission testing problem is the question of peak versus "quasi-peak" (QP) measurements. Whereas the peak detector is ideal for observing lowrepetition-rate impulses which might cause military systems to malfunction (hence, peak

detection is called for in MIL-STD-461 B) , a QP detector's response is proportional to the

subjective annoyance effect experienced by radio broadcast listeners. The Special

International Committee for Radio Interference (CISPR) of the lEC also recommends limits that

have been adopted by many national regulations throughout the world, based on QP measurements.

Figure 9.C.16 shows the relative response of various detectors to an impulse passed through a filter with a -6 dB bandwidth of 9 kHz (as used for conducted emission testing from 150 kHz to

30 MHz) [2]. The CISPR and ANSI detectors are of the QP type (ANSI meter has 600-ms

discharge-time-constant). Recently, NBS has been asked to provide traceability for QP-reading

meters by calibrating the spectral flatness of impulse generators used to align and calibrate

QP meters, particularly those used for emission testing in the military's TEMPEST program

[21].

B. Susceptibility Testing The CEMI problems associated with susceptibility testing are mainly (a) characterizing surge protection devices, and (b) determining the conducted impulse

susceptibility of line operated devices. In both of these kinds of tests, a reliable source

of impulsive waveforms is needed having peak amplitudes in the 1- to 10-kV range with fast

front-edge slopes on the order of 1000 V/ps or faster. Such impulse generators are then used

to simulate the kind of lightning impulse and other waveshapes discussed in section 9.G.1

specified by IEEE Std 587 [7]. Some initial work in characterizing the breakdown voltage of

gas-tube lightning arresters was performed at NBS which relied on the development of a basic

circuit for such testing purposes. Errors in the voltage divider (due to time-constant

differences between the high- and low-side resistors) were determined for the circuit since

these errors are what limit the measurement accuracy in most surge testing applications [22].

Characterizing the properties of surge protection devices has been of major interest, of

course, to the manufacturers of these devices. Numerous application guides, handbooks,

technical tips, etc. are available for describing these products [23]-[25]. These devices

generally fall into two types: crowbar and clamping. Crowbar devices consist mostly of gas

breakdown tubes (or "spark gaps"), carbon-block protectors, and solid-state thyristors.

Crowbar devices provide a blocking voltage which when exceeded (or when the device is

triggered) causes a high-conduction "short" which will continue until the current through the

device is brought back to some low level. Clamping devices, on the other. hand, perform a voltage-limiting function and depend on their nonlinear impedance, acting in conjunction with

the transient source impedance, to limit the surge current. Typical clamping devices consist

of reverse selenium rectifiers, avalanche (zener) diodes, and varistors made of materials such

as silicon carbide, zinc oxide, etc. The difficulty with all of these devices is that they

exhibit properties such as maximum dv/dt or di/dt limits, nonlinear energy absorption ratings,

etc., which make their performance in a given application somewhat unpredictable, unless there

is a significant data base on which to select a design [4], [6], [7] , [ 1 5] , [23].

Conducted impulse susceptibility testing can be carried out in the inverse fashion used

for emission testing. Rather than measuring the emission levels out of the power line cord

from equipment under test (see Figs. 9.C.12 and 9.0.14), an impulsive transient or source of

159
noise is injected into the equipment (either in the common or differential mode) by a generator via a suitable LISN circuit. Commercial surge generators are now available on the

market which provide plug-in modules that permit surge tests to be made conforming to most of

the IEEE, ANSI, lEC, and other telecommunications and automotive engineering surge testing

standards [26]. Nevertheless, there are practical difficulties encountered when trying to

propagate these standard waveforms through various isolating transformers, filters, lines, and

cables [27].

REFERENCES

[1] Showers, R. ; et al . Electromagnetic compatibility in power systems. Tutorial Session,

Power Electronics Specialists Conference; NBS, Gaithersburg, MD; 198^ June l8-2^.

[2] Keiser, B. E. Principles of Electromagnetic Compatibility. Artech House Inc., Revised

Edition, 1983; 57-6M.

[3] Golde, R. H. The frequency of occurrence and the distribution of lightning flashes to transmission lines. AIEE Trans. 6^; 1945.

[4] Martzloff, F. D.; Hahn, G. J. Surge voltage in residential and industrial power

circuits. IEEE Trans. Power Apparatus and Systems; PAS-89(5): 1049-1056; 1970 JulyAugust.

[5] Greenwood, A. Electrical Transients in Power Systems. New York, NY: Wiley

Interscience; 1971.

[6] IEEE Surge Protective Devices Committee, Bibliography on surge voltages in AC power

circuits rated 600 V or less. IEEE Trans. Power Apparatus and Systems, PAS-89(6):

1056-1061; 1970 July-August.

[7] Martzloff, F. D. The development of a guide on surge voltages in low voltage AC power

circuits. Report 8ICRDOM7, General Electric, Schenectady, NY; 1981 (Basis for IEEE

Standard 587-1980)

.

[8] lEC Working Group 28A, Insulation coordination within low-voltage systems, including

clearances and creepage distances for equipment. lEC Report 66^1; 1980.

[9] Fisher, F. A.; Martzloff, F. D. Transient control levels. IEEE Trans. Power Apparatus

and Systems, PAS-95(1); 1976 January-February.

[10] Fisher, F. A.; Martzloff, F. D. Surge Protection Test Handbook, Burlington, MA: KeyTek

Instrument Corp.; 1982.

[11] Bull, J. H. Impedance on the supply mains at radio frequencies. Proc. 1st Symp. Tech.

Exhibition on EMC; 1975 May; Montreux, Switzerland. .

[12] Dixon, D. S. ; et al . Measurement of voltage transients on the 60 Hz powerline of an operational Naval platform. IEEE Symp. on EMC; 198^1 April; San Antonio, TX.

[13] Cannova, S. F. Investigation of short-time voltage transients in shipboard electric

power systems. Lab Project 9^0-55, Progress Report 1, U. S. Naval Applied Science

Laboratory; 1968 September 12.

[14] Cannova, S. F. Interface standard for shipboard systems. DoD-STD-1399 (Navy), Section

300; 1978 August 1

.

[15] Laug, 0. B. ; Hillhouse, D. L. Analysis of requirements for shipboard voltage spike

suppressors. Nat. Bur. Stand. (U.S.) Int. Rept . 84-2855 (For official distribution

only); Gaithersburg, MD; 1984 March.

[16] Laug, 0. B. ; Hillhouse, D. L. Guideline on electrical power for ADP installations.

FIPS PUB 94, 29-51; 1983 September 21.

[17] Vogul , W. The susceptibility of computer systems to noise transients; countermeasures

and test equipment. CIGRE paper published by Study Committee 36 (Interference),

ELECTRA, no. 83.

160
[18] Head, M. Know RF-emission regulations pertaining to your designs. 1M9-15^, EDN ; 1981

August 19.

[19] Mitchell, W. T. FCC and VDE impose tight conducted RFI/EMI specs. 141-147, Electronic

Design; 1982 December 23.

[20] Dash, G. Understanding EMI test methods eases product acceptance. 183-191, EDN; 1983

May 26.

[21] Private communication from J. Durant, Eaton Instruments Division of AILTECH Corp. to B.

A. Bell, Electrosystems Division, NBS.

[22] Hillhouse, D. L. Circuit for impulse testing of gas tube lightning arresters. IEEE

Trans. Communications, C0M-20(5): 936-941; 1972 October.

[23] Hillhouse, D. L. Transient Voltage Suppression Manual, Auburn, NY: 4th edition,

Semiconductor Products Dept., General Electric, 1983.

[2M] Gloer , H. N. Voltage transients and the semiconductor. The Electronic Field Engineer,

2, no. 2; 1979 March/ April.

[25] Clark, 0. M. EMP Transient suppression Using the Transzorb. Tempe, Arizona: General

Semiconductor Industries, Inc., 1979.

[26] Clark, 0. M. Transient testing is spreading to systems designers. Electronic Design;

1976 May 10.

[27] Martzloff , F. D. The propagation and attenuation of surge voltages and surge currents

in low voltage AC circuits. IEEE Trans. Power Apparatus and Systems, PAS- 102, no. 5; 1983 May.

EMITTEB SUSCHPTOR

I SIGNAL OB CONTROL CABLES 4

POWER LINE

A. RADIATION FROM SOURCE CASE TO SUSCEPTOR CASE AND

CABLES (1 and 2) ,

I. RADIATION FROM SOURCE CABLES (ESPECIALLY THE POWER

CABLE) TO SUSCEPTOR CASE AND CABLES (3 and 4) .

C. DIRECT CONDUCTION FROM SOURCE TO SUSCEPTOR VIA

A COMMON CONDUCTOR, FOR EXAMPLE, THE POWER

LINE (S).

Figure 9.C.1 Basic interference coupling mechanisms (from [1])

161
Figure 9.C.2 Ground line and reactance-coupled CEMI

,

10.000

1.000

100^

FREQUENCY

OF VOLTAGE ia

TRANSIENT/YEAR

1.0

0.1

0.01

L

MEASURED DATA U.S. COMMERCIAL BUILDING

fOR 24 HOURS ONLY

THESE ARE THE NEW DATA POINTS

SERVICE ENTRANCE OF BANK

BUILDING iNBASEL.Switzerland

SERVICE ENTRANCE. 16 FAMILY

HOUSE. UNDERGROUND SYSTEM

(HEAD STATION!

FAM HOUSE SUPPLIED BY OVERHEAD

TRANSMISSION LINES. LANDIS & GYR. PLANT. 2UG. OUTLET

IN LAB.

16 FAMILY HOUSE. UPSTAIRS LIVING ROOM OUTLET.

LANDIS & GYR. ZUG. OUTLET IN FURNACE ROOM.

(Courtesy of L. Regez, Landis S Gyr., Zug. Swiuerland)

A220V\

'service

1 .2 .4 .6 1 2 4 6 10 kV

PEAK VALUE OF VOLTAGE TRANSIENT

U.S. COMPOSITE CURVE. (120V)

SERVICE

Figure 9.C.3 Frequency of occurrence of transient overvoltages In 220-V

and 120-V systems (from fl]).

162
isn ^^Mi^^^^^^5S^

un ^^^ ^^^^^^^^ sn r^^^ ^^^^^^^^ VOLTS m

C?-^*"^

^^^^^^^

snt)

inos

ISO ^s

f^^ ^^^^^^^^

SI in

RICIOSECORtS Figure Al. Trmaslent recorded doiinf atartiBf

of a fomaee blower at aerriea box

I.S 53 100 RtCIOSECDIOS

Figure A3. Transient recorded during

unidentified disturbance at service box

XV 75 100 RICtOSCCOROS

S3 ID

Kicioseco.ios Flfvre A2. Transient recorded during U{htnin( storm Flfure A4. Composite recording of furnace Ignition on street pole transformer transients over 24 hours at service box

Figure 9.C.4 Typical waveshapes of ac power line transients

(from [9]).

163
CATEGORY A

CATEGORY B

•kV, 100kHz SOOA

•kV, 100UU SOOA AND

tkVK.

U s SO >3kA,lx20

^

y^

OUTSIDE AND

s^ SERVICE ENTRANCE

CATEGORY A . CATEGORY B \

OUTLETS a LONG ' MAJOR FEEDERS a\

BRANCH CIRCUITS SHORT BRANCH

CIRCUITS

Figure 9.c.f IEEE Std. 587 test waveforms relative to the

ac power feeder line (from [7] and [10]).

164
•00 1200 1600 PEAK AMPLITUDE. VOLTS

Figure 9.C.6 Seven-day histogram of shipboard transieni

voltages in 120- and 440-V systems (from

[13] and [14]).

165
NUMBER OF OCCURRENCES

85 95 1 05

PEAK VOLTAGE

125 135 145

(a)

NUMBER OF OCCURRENCES

Figure 9.c.7

84 100 PEAK VOLTAGE

(b)

75-hcur histograms of voltage transients recorded on the

450-V, 60-Hz power panel of an operational Naval platform,

(a) measurement system //I, (b) measurement system //2 (from [12]) .

166
CLAMPING L/V£i.

CLAMPlSib LeV£L

Figure 9.C.8 Steady-state power voltage wave (60 or 400 Hz)

with superimposed transients, showing the effect

of clamping-type suppressors on such transients

(a) when the clamping level is set correctly; and

(b) when the clamping level is set too low. ([15]).

167
/

y iUPPR£S&OfK

Figure 9.C.9. Circuit for calculating the current and energy

dissipated in a clamping suppressor (from [15])

FLOATING GROUND IWL HE HE

SINGLE-POINT GROUND

HULTIPLE-POINT GROUND T JT I

Figure 9.C.10 Fundamental grounding concepts (from [2])

168
Stnqf po*nr ftiiy >titci iQf pOw*«. iMvcommu/^caiiont *nd W«.

mI«iv can b« *ppfo«crv»<i Dy: (a) punmg p«n«u on um« mm of foom. 101 t«OAfattf%g xhtn no moi« Uun 2 lO 4 (••t. and (c) tftisf- COnnAcimQ in«w ancio&urM win conouclmg ttr*o«- ThM wril •Qw«M<« mofti ground voiUQ* dtftaranca* 0«r««««n tf>«m up to

1 MHf Of mor«.

Figure 9.C.11 Electrical interface location for an ADP facility

(from [16]).

169
EQUIPMENT UNDER TEST

POWER LINE CORDv^

H I 1 i^.'* - _

CABLES

Figure 9.C.12. Basic conducted emission test conditions controlled by FCC

(parts 15 and 18) Rules.

100

IMi>£OANCt

10

TOLERANCE 420%

bCSlCN CHAMACKAlSTlC

IOLEIUMCC iNH

•.I 14 »MOMMC>iUH(|

J 1 I I I I I I -I I I I I I I -L. I I I I I I

0.19

S < 7 •«!

1.0

4 S t > • 9 I 10.0

1 4 S 6 > •»

I

30.0 MMZ

FKeOUCNCY

Figure 9.C.13 Line-impedance characteristic (inset) for making conducted

emission measurements, as mandated by FCC Docket 20780

(from [1] and [18]).

170
<u

CO •Ho

c

T5

0) 4JU

3

•Vc

o

uc

•Ho

2 /^

M ON

> o

r-l «4-l

I ^^ a

o c

in (UE

O -H3

0) cr

i-l (U •H

CO U^

AJ o

0) •O COc u O

•H -H 3 CO O CO U 1-1 •H E

ON

3

00

To power source

171
100-

90-

80-

> 70-

m

60-

so40-

CONDUCTED EMISSION LIMITS

FCC - VDE

N N

S

N

s \

^PROPOSED N

P.C.C. CLASS A s

S s VDE 087IA

^x

1 Klcf^'^

CA

FCC. CLASS 9 5M\^

1 1

1 1 0.1 1.0 FREQUENCY (MHz)

10 100

Figure 9.C.15 Conducted emission limits set by the FCC and VDE

(from [19]) .

1 iia

«i«

*H5I

ClSPt "^

^^ ^

^^^.^ TXUC (as ^<^

^

1 ^ ? '° "i^ ^ ' ( U KX

IMMUC KKTITKII UTC (Mil

Figure 9.C.16 Relative resDonse of various detectors

to an impulse passed through a filter

with a -6 dB bandwidth of 9 kHz

(from [2]).

•ttU.S. GOVERNMENT PRINTING OFFICE: 1986 672-641/45286
NBS'lUA (REV. 2-8C1 U.S. DEPT. OF COMM. BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NBS/TN-1099

2. Performing Organ. Report No, 3. Publication Date

July 1986

4. TITLE AND SUBTITLE

Electromagnetic Compatibility and Interference Metrology

5. AUTHOR(S)

M. T. Ma and M. Kanda

6. PERFORMING ORGANIZATION (If joint or other than NBS. see instructions)

NATIONAL BUREAU OF STANDARDS

DEPARTMENT OF COMMERCE

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State, ZIP)

10. SUPPLEMENTARY NOTES

I I Document describes a computer program; SF-i85, FlPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less factual sumnnary of most si gnificant information. If document includes a significant

bibliography or literature survey, mention it here) The material included in this report is intended for a short course on electromagnetic

compatibility/interference (EMC/EMI) metrology to be offered jointly by the staff of the

Fields Characterization Group (723.03) and the Interference Characterization Group

(723.04) of the Electromagnetic Fields Division (723). The purpose of this short course

is to present a review of some of the radiated EMC/EMI measurement methods, to which the

National Bureau of Standards (NBS) at Boulder, Colorado, has made significant contributions during the past two decades. The technical foundation for these methods, and

interpretations of the measured results are emphasized, as well as strengths and limitations. The entire course is presented in nine chapters with the introductory part given

as Chapter 1. The particular measurement topics to be covered are: i) open sites

(Chapters 2 and 6) , ii) transverse electromagnetic cells (Chapter 3) , iii) techniques

for measuring the electromagnetic shielding of materials (Chapter 4) , iv) anechoic

chambers (Chapter 5), and v) reverberating chambers (Chapter 8). In addition, since

small probe antennas play an important role in some of the EMC/EMI measurements covered

herein, a separate chapter on various probe systems developed at NBS is given in Chapter

7. Selected contemporary EMI topics such as the characterization and measurement of a complex EM environment, interferences in the form of out-of-band receptions to an antenna, and some conducted EMI problems are also briefly discussed (Chapter 9).

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

anechoic chamber; complex electromagnetic environment; conducted electromagnetic interference (CEMI) ; electromagnetic compatibility (EMC); measurement methodology; open sites;

out-of-band responses; probes; radiated electromagnetic interference (EMI); reverberating

chamber; shielding of materials; transverse electromagnetic (TEM) cell.

13. AVAILABILITY

I^Unllmited

I I For Official Distribution. Do Not Release to NTIS

r^f-Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C. 20402.

[2Z\ Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

178

15. Price

USCOMM-DC 6043-P80

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