A 11 ID 3 ISITD?

NATL INST OF STANDARDS & TECH R.I.C. T

A1 11 031 51 907 -,«»,«

Daywltt, William C/Radlometer equation a TION5

QC100 .U5753 N0.1327 1989 V198 C.1 NISTSrATES 0* */

NIST TECHNICAL NOTE 1327

U.S. DEPARTMENT OF COMMERCE / National Institute of Standards and Technology
NATIONAL INSTITUTE OF STANDARDS &

TECHNOLOGY

Research Information Center

Gakhersburg, MD 20899
delbo

.MOS3

Radiometer Equation and : ^ Analysis of Systematic Errors

for the NIST Automated Radiometers

William C. Daywitt

Electromagnetic Fields Division

Center for Electronics and Electrical Engineering

National Engineering Laboratory

National Institute of Standards and Technology

Boulder, Colorado 80303-3328

U.S. DEPARTMENT OF COMMERCE, Robert A. Mosbacher, Secretary

Ernest Ambler, Acting Under Secretary for Technology

NATIONAL INSTITUTE OF STANDARDS AND TECHNOLOGY, Raymond G. Kammer, Acting Director

Issued March 1989
National Institute of Standards and Technology Technical Note 1327

Natl. Inst. Stand. Technol., Tech Note 1327, 28 pages (Mar. 1989)

CODEN:NTNOEF

U.S. GOVERNMENT PRINTING OFFICE

WASHINGTON: 1989

For sale by the Superintendent of Documents, U.S. Government Printing Office, Washington, DC 20402-9325
CONTENTS

Page

1

.

Introduction 1

2. Radiometer Equation 2

3. Cryogenic Standard Error 5

4. Ambient Standard Error 5

5. Power Ratio Error 6

6. Mismatch Factor Error 6

7. Asymmetry Measurement and Error 7

8. Isolation Error 10

9. Connector Error 12

10. Frequency Offset Error 14

11. Nonlinearity Error 14

12. Total Error 17

13. Summary, Discussion, and Conclusions 18

14. Acknowledgments 19

15. References 21

in

Radiometer Equation and Analysis of Systematic Errors for the NIST Automated Radiometers

W.C. Daywitt

Equations used in the NIST coaxial and waveguide automated radiometers to estimate the noise

temperature and associated errors of a single-port noise source are derived in this report. These

equations form the foundation upon which the microwave and milhmeterwave noise calibration and

special test services are performed. Results from the 1-12 GHz coaxial radiometer are presented.

Key words: calibration; error analysis; microwave; millimeterwave; noise temperature; radiometer.

1. Introduction

The Electromagnetic Fields Division of the National Institute of Standards and Technology (NIST) has

two types of radiometers in operation at the present time, the switching and total-power radiometers. The

radiometer and error analysis equations for the WR90 and WR62 switching radiometers have been previously

described [l]. Similar equations for the automated, total-power radiometers are derived in this report. These

radiometers are designed to provide noise temperature calibrations and special test services for coaxial and

waveguide noise sources. The coaxial radiometer operates from 1 to 12 GHz in the following frequency

ranges: 350 MHz to 500 MHz, 500 MHz to 1 GHz, 1 to 2 GHz, 2 to 4 GHz, 4 to 8 GHz, and 8 to 12 GHz;

a separate r.f. front end being supplied for each of these ranges. Results from the 2 to 4 GHz range will be

presented in this report. The hardware and software aspects of the coaxial radiometer is presented in [2].

The milhmeterwave radiometers are under construction, and consist of the WR10, WR15, WR22, WR28,

and WR42 waveguide bands.

A simplified schematic diagram of the system is shown in figure 1, where the directional coupler, isolator,

r.f. amplifier, and mixer represent any one of the front ends just mentioned, while the 6-port circuitry is common to all frequency ranges. The system performs two functions. With the switch in the 'up' position

as shown in the diagram it operates as a radiometer with the measurement sequence determining both the

effective input noise temperature [3] Te of the system and the noise temperature Tx of the DUT (device

under test) by comparison against the ambient Ta and cryogenic Ts noise temperature standards. With the switch in the 'down' position it operates as a single 6-port, vector reflectometer for determining the complex

reflection coefficients of the DUT, the standards, and the input port of the system. In the radiometer mode,

switches at the system input (not shown in the diagram) alternately sample the noise temperatures T„

,

Ta , and Tx at which time their respective down-converted, i.f. powers are determined by the detector. An

estimate of Tx is made by inserting ratios of these powers and the reflection coefficients into a radiometer

equation. A derivation of this radiometer equation along with the associated errors is the subject of this report, and is based upon analytical tools to be found elsewhere in the literature [4], [5].
2. Radiometer Equation

A theoretical examination of the measurement technique leads to an equation relating 1) the various power

and reflection coefficient measurements made with the system to 2) the noise temperature of the DUT and

the two noise standards. When this equation is solved for the noise temperature Tx of the DUT there results an equation which for convenience is called the radiometer equation. Figure 2 is a schematic diagram of that portion of the system needed for the analysis, consisting of a front end between the noise sources Tx and T,, and the switch at the receiver and 6-port inputs. This switch following the isolator is set to the

receiver position for the noise and asymmetry measurements, or to the 6-port reflectometer for the reflection measurements. The front end switches (22 through 25 in the diagram) correspond to the identically numbered

switches in figure 3 of the operating manual [2]. The net i.f. powers Px , Pay and Ps detected by the power

meter are related to the r.f. spectral powers [3] px , pa , and p, by the equation

Pi = gBpi {i = x,a,s) (2.1)

where g is the system gain at the measurement frequency and B is the system noise (or convolution [6]) bandwidth. The gain-bandwidth product in (2.1) will be dropped since only power ratios are needed in the

following analysis, and Boltzmann's constant is set equal to unity in the following so that spectral powers

and noise temperatures may be used interchangably. When the DUT is connected to the system input at switch 25, and switches 22 and 23 are in their 'up' positions, the spectral power delivered to the receiver is

px = Mx Tx r, x + Ta {Nx - Mx r, x ) + Nx Te (2.2)

where Mx and Nx are mismatch factors [4] at the ports indicated in the figure for different switch positions,

r\ x is the efficiency of the front end from port 3 to the isolator output, Ta is the ambient temperature of the

switches, and Te is the effective input noise temperature of the receiver and power meter combination. The

second term in (2.2) accounts for the thermal noise generated in the switches and the isolator.

When switch 22 is in the 'down' position, the power is

Pa = Na Ta + Na Te

. (2.3)

With switches 23 and 24 'down' and switch 22 'up' the power is

p, = Ms T,r, B + Ta {Nt -M. V.) + Na Te

. (2.4)

The following ratios are calculated from the measured powers:

Y. = px /pa (2.5)

and

Y,=ps /Pa . (2.6)

Equations (2.3) through (2.6) can then be combined and lead to Tx = Ta + (M.!j./MxiJx)[(Yx - 1)/(Y. - 1)](T. - T.) + (ATx ) ISO (2.7)

2
CRYOGENIC

STANDARD —

©

AMBIENT f—

STANDARD

IMPEDANCE

STANDARDS —

©

SYSTEM

INPUT

PLANE

Figure 1. A simplified schematic diagram of the automated radiometer.

DUT

CRYOGENIC

STANDARD

'UP* 'DOWN'

AMBIENT

STANDARD

I "1 NJe . N aTe , NsTe Nx

, N a, Ns

2x,2a,2s

RADIOMETER

RECEIVER

(g,B)

6— PORT

REFLECTOMETER

NBS

TYPE IV POWER METER

IF

Pa

•Ps

Figure 2. A schematic diagram of the automated radiometer's r.f. front end.
where the last term is an uncertainty due to the finite isolation of the isolator in figure 2. This error is a result

of the different impedances seen by the receiver as the system samples power from the three noise sources. This changing impedance affects both the spectral powers and the value of the receiver noise temperature

as reflected in the equation

(Ar.)/so = (ATX ) N + (ATx ) e (2.8)

where

(ATX ) N = {Ta /Mx r, x )[(Yx - 1){N3 - Na )/(YS - 1) + Na - Nx ] (2.9)

and

(AT*), = (l/Mx r, x )[(Yx - 1)/(Y, - l)}(N,Te - Na Te ). (2.10)

The boldfaced portion of (2.7) (i.e. without the isolator error term) is the radiometer equation that relates the

DUT noise temperature Tx to the ambient and cryogenic noise temperatures (Ta and T9 ) and the measured

Y-factor ratios and mismatch factors. More explicit expressions for the M's, r?'s, {ATx )iso, T, and Ta will be developed later.

Although the effective input noise temperature Te and the system gain g are not needed in the radiometer

equation, these parameters are monitored to help validate system performance. The Y-factor technique [7] is used to obtain a rough estimate of the effective input noise temperature of the system at the receiver input in figure 2 by switching between the cryogenic and ambient noise standards. The derivation is straightforward

and leads to re = (ra -yr,)/(F-i) (2.11)

where Y is the measured ratio of: 1) the spectral power with the ambient source connected to the system

input; to 2) the spectral power with the cryogenic source connected to the system. The front-end switches

are assumed to be lossless and reflectionless for this estimate.

An approximate value for the system gain is also of interest and can be obtained from

g = Pa /k{Ta + Te )B (2.12)

where Pa is the power at the power meter with switch 22 in the 'down' position, and A; is Boltzmann's

constant. The noise bandwidth B is measured before the radiometer measurement, and is stored in the

system software for use in (2.12).
3. Cryogenic Standard Error

The DUT noise temperature error 6TX due to an uncertainty 6Tt in the cryogenic standard is found by

differentiating the radiometer equation with respect to Ta , multiplying the result by 6Tt , and discarding

Ma /Mx and rj s /rix since these two factors are close to unity. The corresponding relative error £a in the DUT

noise temperature is

E. = 6TX /TX = (Ts /Tx ) | (Tx - Ta )/(T, - Ta ) \ (6T,/TS ). (3.1)

The last factor is the relative uncertainty in the standard noise temperature, and takes a different form for the coaxial [8a] and waveguide [8b] noise standards.

The relative noise temperature uncertainty of the coaxial standard for frequencies between 1 and 12.4 GHz is [8a] 6T>I T> = (^29} + K-o!^ ± °-°5

' T' l]/1/2 ± °-°036/ % (32)

where / is the frequency in GHz, and r„ is the reflection coefficient of the standard.

4. Ambient Standard Error

Ignoring the mismatch factors and asymmetry ratio in the radiometer equation shows that an uncertainty

of STa in the ambient standard output leads to a relative error £a of

£a = (Ta /Tx ) I 1 - (J* - Ta )/{T, - Ta ) | {6Ta /Ta ) (4.1)

where 6Ta /Ta is the relative uncertainty in the ambient standard output. With an assumed value of ±0.25 K

for 6Ta and a nominal value of 297 K for Ta ,

6Ta /Ta = 0.17 %. (4.2)
5. Power Ratio Error

By ignoring the mismatch factors and the asymmetry ratio, the first-order differential of the radiometer

equation with respect to the Y ratios leads to a relative error £y in the DUT noise temperature equal to

€y =| 1 - Ta /Tx I 6[(YX - 1)/(Y, - 1)}/[(YX - 1)/(Y, - 1)]. (5.1)

Previous tests [9] have shown that the NBS Type IV power bridge [10] is capable of measuring power

ratios to better than ±0.002 dB, and that the error is random and independent of the value of the ratio. The test described in reference [9] was repeated (using a non-precision attenuator) on the present radiometer

with the 2-4 GHz front end, and resulted in an error of ±0.015 dB (see figure 3). Although it is thought

that most of the error appearing in figure 3 is due to the non-precision r.f. attenuator, for the present an

error of ±0.01 dB for the uncertainty of the ratio in (5.1) is assumed. This results in a 0.23% error in the

last factor of (5.1), and an error in the DUT temperature of

ly = 0.23 | 1 - Ta /Tx | %. (5.2)

6. Mismatch Factor Error

Approximate expressions for the mismatch factors Mx and M„ can be obtained by using figure 2 and

expressing the complex reflection coefficients appearing there in terms of their real (x) and imaginary (y)

parts:

M.^i-ir^Ki-iran/li-r^i2

• 1 ( \* ( x \* { ] = 1 - [xx - x3 ) - [yx + y3 ) and

M. = (i-|r. | 2 )(i-|ro|2 )/|i-r8 r | 2 = 1 - [x, - so) - (ys + yo) • Combining these equations leads to Mt /Mx = 1 - (xt - x ) 2 - (ya + y ) 2 + {xx - x3 ) 2 + (yx + y3 ) 2

. (6.3) The relative first-order uncertainty in this ratio is 6[{M,/MX )]/{M,/MX ) = 2[(s. - x3 ) (6xx - Sx3) - (x, - x ) (6xs - 6x ) , ^ (6.4) + {yx + y3){hx + h3) - (y3 + yo)(<$y« + ho)}-

Numerous 6-port reflectometer measurements at 94 GHz in the WR10 waveguide band and in the 2-4 GHz frequency range indicate that the systematic errors in x and y are less than ±0.005. These values are assumed to apply in the present case until a more complete error analysis of the 6-port measurements can be

performed. Assuming the systematic errors to be independent of the magnitude of the real and imaginary

parts of the reflection coefficients, only the last two terms of (6.4) survive, leading to 6[{M./Mx )]/{M./Mx )=\yx +y3 -y.-yo\ %. (6.5)

The form of the radiometer equation shows that the relative error Sm in Tx due to mismatch uncertainty

is given by

em = S[(M,/Mx )\/(M9 /Mx ). (6.6)

6
NOISE^

SOURCE

0-^-0-

A

NBS TYPE IV POWER

METER

CQ

E

*0

15

10

5 —

-5

-10

-15

H—HB

I

H 1 H 10

A(dB)-

15

A = 10 log

Yi-1

Yo-1

Figure 3. The noise source at the top of the figure terminates a precision variable-vane r.f. attenuator. Two

Y factors, Y\ and Y2, are determined by the Type-IV power meter for two different settings of the attenuator corresponding to an attenuation difference of A in dB. If the attenuator and power

meter are error free, these quantities are related by the equation at the bottom of the figure. The

actual error £ is the difference between the left and right hand members of this equation, and is plotted in mdB along the ordinate.

7. Asymmetry Measurement and Error

The presence of the efficiency ratio rj s /rjx in (2.7) is due to the asymmetry in path efficiencies between the

path from port 3 (see fig 2) to the receiver and the path from port to the receiver. The measurement of

this asymmetry is performed by using two solid-state noise diodes in the setup shown in figures 4a and 4b,

which are simplified schematic diagrams of the portion of the system preceding the receiver in figure 2. With

diode No. 1 (Tx i) connected to port 3 (fig 4a) and switches 22, 23, and 25 in the 'up' position, the power
delivered to the receiver is

Pxi =Mxl ri x {Txl -Ta ) + NTa + NTe (7.1) where Mx \ and N are mismatch factors at the ports shown, Ta is the ambient temperature of the two switches

and the isolator, and NTe

is the effective input noise temperature of the receiver. Sufficient isolation is present

to insure that the mismatch factor at the isolator-receiver junction remains constant, independent of the

switch positions. With switches 23 and 24 'down'

Px2 = Mx2 r,,{Tx2 - Ta ) + NTa + NTe (7.2)

and with switch 22 'down'

pa = NTa + NTe

. (7.3)

Equations (7.1), (7.2), and (7.3) can be combined to give

V.fa* = (Mxl /Mx2 )[Txl - Ta )/{Tx2 - Ta )}{Yx2 - l)/(Yxl - 1) (7.4)

where Yx\ is the measured ratio of px2 to pa , and Yx2 is the ratio of px \ to pa .

The two noise diodes are interchanged and the process repeated (fig 4b), yielding

*./f. = iKa/KiW^ - Ta )/(Txl - Ta )){YU - 1)/(FX

'2 - 1) (7.5)

where Yxl is the ratio of p'xl to pa , and Yx2 is the ratio of p'x2 to pa .

After multiplying (7.4) and (7.5) together and taking the square root,

*/*. = {[Mxl M'x2 /Mx2 M'xl \{Yx2 - \){Y'xl - l)/[(Fxl - \){Y'x2 - I)]} 1/*. (7.6)

Equation (7.6) is used to determine the path-asymmetry ratio, with the mismatch factors determined by

6-port reflectometer measurements.

The relative error £at in the DUT noise temperature can be estimated by taking the first-order differ- ential of (7.6) with respect to the asymmetry ratio, and leads to

£at =| 1 - Ta /Tx | 6(r)„/rix ) = 0.23 | 1 - Ta /Tx | 6 A %

where <5A is the dB uncertainty in the asymmetry ratio.

A (see fig 3) has only been estimated for the 2-4 GHz band, resulting in a value of ± 0.01 dB. This value was determined from the following three pieces of information: 1) when the asymmetry ratio was measured

at 100 MHz steps across the 2-4 GHz frequency range it was found to lie within a ± 0.01 dB spread (r),/rj x within the range from 0.998 to 1.002) and to be randomly distributed; 2) when the attenuation of a 3 dB

pad was measured by performing an asymmetry measurement with and without the pad at one of the ports

its value was within 0.01 dB of the value measured by a dual 6-port ANA; and 3) when a noise diode was calibrated with and without the 3 dB pad between the diode and the system input, the resulting noise

temperatures (~ 10000 K) agreed to within 10 K. If this entire 10 K difference were attributed to asymmetry

ratio uncertainty the uncertainty would be less than 0.004 dB. Therefore, (7.7) with a (5 A) of ±0.01 dB for the 2-4 GHz frequency range appears to be safe and leads to

£at = 0.23 | 1 - TJTX | %. (7.8)
a)

M'

/

t ; x2

,/ PORT 3

Tv oV/ XSW25

1>

UP

*1

N

Px2"

Pa'

/

SW22 > - Ta

v T. P

NT,

«~r

/

/

/

/

b)

Figures 4. A simplified schematic diagram of the radiometer front end with a) diode 1 at port 3 and diode 2

at port 0; and with b) diode 2 at port 3 and diode 1 at port 0.
8. Isolation Error

The impedance seen by the receiver changes slightly due to the finite isolation of the isolator in figure 2 as the noise sources Tx , Ta and T„ are connected to the system. The receiver noise contributions Nx Te , Na Te , and Nt Te vary correspondingly. The resulting error in the DUT noise temperature is calculated from (2.8),

(2.9) and (2.10). The error due to a changing mismatch factor, (2.9), is estimated first.

The mismatch factor at the isolator-receiver connector for a particular switch arrangement where source

i (i=x, a,or s) is connected to the receiver is

Ni = (l- | r2i | 2 )(i- | rr | 2 )/ 1 i - r2i vr | 2

. (s.i)

The reflection coefficient looking back into the isolator is approximated by the equation

r2j = S22 + S12S21T1 (8.2)

where the S's are the scattering coefficients of the switch-isolator combination which are assumed to be

independent of switch position. The squared magnitude of this equation is approximately

I r2, | 2 =| S22 1 2 +2ReS2

*2 s12 s21 ri (8.3)

where the asterisk indicates the complex conjugate. Equation (8.1) shows that the change in N when source

i is replaced by source j is ANi3- = Ni- N,-

(8-4) =|r2y|

2 -|r2i | 2 +2Rerr (r2i -r2y ).

In terms of (8.2) and (8.3), (8.4) becomes

ANij = 2ReS12 521 (rr - S*2 )(I\ - ry ). (8.5)

Using this equation in equation (2.9) yields the approximation

(ATX ) N = 2Ta [(Tx - Ta )/(T, - Ta )} ReS12 S21 (I\. - S2

*2 )(I\ - Ta ) + Res12 s21 (rr - 5*

2)(ra - rx ).

Before reducing this error further it will be combined with the error due to changing receiver noise to be

estimated next.

The effective temperature (power) delivered to the receiver input which accounts for the noise generated

in the receiver itself can be expressed in the form [11] NTe = T1 +T2 \T2-i* | 2 (8.7)

where Ti and T2 are effective temperatures and where 7* is a complex constant. Each of these are determined

experimentally. T2 represents any of the reflection coefficients r2a;) T2a , or T2s . With a change from source

i to source j, ANTeij = (NTe )

{ - (NTe ) 3

- = r2 (|r2i - 7

*| 2 -|r2y - 7*| 2

) (8.8) = T2 [\ r2t-

| 2 - | Y2j | 2 -2Re7*(72,' - %/)]•

10
Using (8.2) in the last equation gives

ANTeij = 2T2 ReSi2 S21 (S;2 - 1*)(Ti - T3 ) (8.9)

and the corresponding DUT error becomes

[ATx ) e = 2T2 [((TX - Ta )/(T, - Ta )) ReS12 S21 (S;2 - 1 *)(T t - Ta ) + Re512 521 (5*2 - 7*)(ra -rx )].

(8.10)

Combining (8.6) and (8.10) leads to

(ATx ) ISO = 2Re{[(T2 - Ta )S*22 + Ta Tr - T2l *\

\{TX - Ta )Tt /{T, - Ta ) + (Tt - Tx )Ta/{T, - Ta ) - Yx \).

(8.11)

Noting that | S21 \ is approximately 1, and taking the maximum possible value of the right-hand side of

(8.11) leads to

(AT^/so = 2 | S12 | {[| S22 || T2 -Ta \ +Ta | Tr | +T2 | 7

*

|]

[| (Tx - Ta )/(T, - Ta ) || T. | + | {Tx - Ta )/{T, - Ta)\\ Ta \ + \ Tx |]}.

Previous measurements [ll] and the fact that the effective input noise temperature of the automated ra- diometer is approximately 1000 K lead to the following values : T2 = 500 K and | 7* |= 0.25. The assumed

isolator parameters are | S2 i |~ 0.003 (corresponding to three isolators with a total minimum isolation of 50 dB) and | S22 \= 0.1. A value of 0.1 for | Tr | and | Ta \, 300 K for Ta , and 77 K for Tt are also assumed.

With these values and (8.12), the relative error & in the DUT noise temperature due to finite isolation is

d = {ATx ) ISO /Tx = 0.47 I T. || 1 - Ta /Tx | +0.047 | 1 - T,/Tx \ +100 | Tx \ /Tx %. (8.13)

11
Figure 5. A schematic diagram indicating the connector efficiencies rj xc , rj ac , and r\ ac at the DUT, cryogenic,

and ambient ports respectively.

9. Connector Error

Ideal connectors at the system input ports have been assumed in the development of the radiometer equation.

To insure low loss and high repeatability at these ports, the highest quality connectors are used on the system

and on the ambient and cryogenic standards. To see how imperfect connectors affect the noise measurement

consider figure 5, where r] xc , rj ac , and rj tc are the connector efficiencies at the DUT, ambient, and cryogenic

standard ports respectively. A development similar to that of section 2 reveals a modified radiometer equation

of the form

Tx = Ta + {Mt /Mx ){r, s lr, x )\{Yx - 1)/{Y. - l)](T, - Ta )( Vsc /r, sx ). (9.1)

This differs from the radiometer equation (2.7) for ideal connectors by the presence of the last factor r\ iC jr\ cx . The efficiency of the ambient termination connector pair is absent from this equation because the switches

and this termination are at the same temperature. Comparison of (9.1) and (2.7) leads to the following

expression for the relative DUT noise temperature error due to ignoring the last factor in (9.1):

£c =| Igc/Vxc — 1 (9.2)

This error consists of a systematic part which accounts for the fact that the DUT connector may have

a different metallic conductivity or contact resistance than the cryogenic standard, and a random part

accounting for connector non-repeatability. The non-repeatability component is included with the systematic

component because the sources remain connected to the system throughout a complete calibration procedure.

Connector repeatability is an incompletely explored subject at the present time so a final analysis of

12
non-repeatability error is not possible. Therefore, what follows is a temporary estimate to be updated when

a better treatment is available. Thirteen sets of measurements of the DUT noise temperature were made at

3 GHz with the standard and DUT noise sources being removed and reattached to the system between each

set, each set containing 100 DUT noise temperature determinations. The standard deviation of the mean

was estimated for the 100 points of each of the 13 sets, the average of which was 15.7 K. This is the amount

of inherent random fluctuation to be expected. The standard deviation of the 13 data points was 24.4 K. Assuming that 15.7 K of this value is due to the inherent fluctuations, the remaining 8.7 K (24.4-15.7) is due to connector non-repeatability. This value is 0.087% of the average DUT noise temperature at 3 GHz.

The value appropriate for other frequencies is obtained by multiplying by (//3) ' since connector loss is assumed to vary as the square root of the frequency. Therefore, the relative DUT noise temperature error due to connector non-repe at ability is 0.050/ 1 ' 2 % where / is the frequency in GHz.

The maximum relative DUT noise temperature error due to the DUT and system conductivities being

different is given by the expression

0.035/ 1 / 2

| 1 - Ta/Tx | % (9.3)

where / is the frequency in GHz. Combining this error with the non-repeatability error leads to

£c = 0.050/1 / 2 + 0.035/1 / 2

| 1 - Ta /Tx | %. (9.4)

The noise temperatures for the ambient and cryogenic noise standards, Ta and TB , are referenced to the

planes immediately preceding their respective connectors. As a result, the connector losses are included as part of the radiometer during the DUT noise temperature measurement as is the DUT connector loss when

the DUT is connected to the system. Therefore, the DUT noise temperature Tx is referred to a plane just

preceding the DUT connector, and does not include the connector noise contribution. This way of defining

the reference planes is the most useful since connector losses are variable and usually ignored in practice.

13
10. Frequency Offset Error

The noise measurement system employs a superheterodyne receiver which, due to the broad-spectrum character of the noise signal, selects upper and lower sidebands around the local oscillator (LO) frequency for processing. In addition, the efficiency ratio (see (2.7)) is also determined in this mode of operation. The

mismatch factor ratio, however, is determined by operating the system in its reflectometer mode where the

various reflection coefficients making up the mismatch factors (see (6.1) and (6.2)) are determined at the

calibration frequency, not the LO sideband frequencies. This ordinarily causes no measurable error unless

there is a long transmission line between the system input ports and the isolator in figure 2. If this length

is denoted by £ (the length from port or port 3 to the isolator) the relative error C in the DUT noise

temperature is £ = 200 I cos(4tt/£/30) sinc(*-££/l5) - 1 I (I I\ I + I Tx \) (10.1) |«u||i-ra /r,| %

where / is the i.f. frequency in GHz, I is the length in centimeters, B is the i.f. frequency bandwidth in GHz, "sine" stands for sin(x)/i, and 8n is the isolator input reflection coefficient. For a /,£,£?, and | su \ of 0.03 GHz, 15 cm, 0.01 GHz, and 0.1 respectively

6, = o.36(| r. | + | r. |) 1 1 - lyr, | %. (10.2)

11. Nonlinearity Error

There are three nonlinearities in the radiometer: mixer nonlinearity; amplifier compression; and nonlinearities

in the thermistor mount used to detect the radiometer output power. Mixer nonlinearity refers to a mixer

i.f. output which contains exponents of the r.f. signal (the signal in the present case is the wave amplitude

from a noise source connected to the input of the radiometer) greater than the first power. A simple analysis

[12] shows that the next exponent to appear after the first in the i.f. output can be no less than the third

power . Assuming this exponent to be present, its coefficient would be proportional to (Ps /Pio) 3 ' 2 where

Ps and Pi are the signal and local oscillator powers respectively. For a noise source with a 104 K noise

temperature, a local oscillator power of 1 mW, and a system noise bandwidth of 107 Hz, this coefficient is of the order of 10-14

. The resulting error in the measured DUT noise temperature is insignificant.

The DUT noise temperature error caused by amplifier compression and power meter nonlinearities are accounted for by treating both together as a variable gain (a gain which is a function of an input power). The

resulting nonlinearity constant aBG (to be defined in a moment) is a measure of this system nonlinearity

and is carefully determined for the system before any noise calibrations are performed, and then carried in computer memory for later error calculations. Figure 6 is a schematic diagram of the model used to develop

the error equation, and consists of three noise sources, an r.f. amplifier and mixer, a local oscillator, an i.f. attenuator and amplifier, and a power meter. The noise temperatures of the DUT, ambient, and cryogenic

sources are Tx , Ta , and T, respectively. The r.f. and i.f. amplifiers have effective input noise temperatures

of Te i and Te2 , and the gain of the r.f. amplifier is g\. The i.f. amplifier gain is variable and assumes the

form 32/(1 + a-QilIk}, where 32 is the gain of a fictitious amplifier in which the constant a vanishes. The

input and output powers of the i.f. amplifier are q and p, and k is Boltzmann's constant. Ta and /? are the

ambient temperature and efficiency of the variable attenuator between the two amplifiers.

14
^sT €> —MV—€) —m—

©

©—

^

v

RF (or 1st IF)

I 1 + ag 2q/k

TV J "V

IF & POWER METER

Figure 6. A schematic diagram of the r.f. and i.f. portions of the automated radiometer.

The input power q is

q = kBgi p{T+Te ) (11.1)

where the system effective input noise temperature referred to the r.f. amplifier input port is

Te = Tel + [T„(l - /?) + Tal/gifi . (11.2)

The gi/3 factor in the denominator of (11.2) remains large enough during the nonlinearity measurement (/?

is changed as part of the measurement process) to assume that Te is constant and equal to Te \. The output

power p detected by the power meter is related to q by the formula in the figure which leads to

where

p = kBG{T + Te )/[1 + aBG{T + Te )\

G = gifig2 .

(11.3)

(11.4)

Denoting the powers with the various noise sources connected to the radiometer input by px , pa , and

p„, and using (11.3) leads to

where

Tx -Ta = (T. - Ta )[(Yx - \)/{Y. - 1)]

• {[1 + aBG{Tx + r.)l/[l + aBG{T, + Te )]}

Yx = px /pa

15

(11.5)

(11.6)
and

Y,=pB /pa (11.7)

are the Y factors determined from the power measurements. Equation (11.5) is an "exact" formula between

the noise temperatures that includes the variable gain of the system. The formula used to estimate the DUT

temperature is f* ~ Ta = (T. - Ta )(Yx - l)/(n - 1) (11.8)

where Tx is the estimated temperature. Combining (11.5) and (11.8) leads to the approximation

Tx =tx + (fx - Ta )(fx - T,)aBG (11.9)

relating the true DUT temperature Tx and the estimated temperature, where the nonlinearity constant aBG is small and accounts for amplifier compression and power meter nonlinearity. Amplifier compression

makes the constant a in this last equation positive while the power meter nonlinearity causes a negative a. Therefore, the product aBG can be positive or negative.

The constant aBG corresponds to an optimum value (a value that gives a minimum for a) of ft deter- mined during the nonlinearity measurement, and is used in subsequent noise temperature measurements.

Once /3 is set and a measurement is made to determine an approximate value for the corresponding DUT

noise temperature T, a value for aBG can be found by making two more measurements of the DUT temperature, 7i and T-2, for two different values for the i.f. attenuator efficiency, fa and fa. The constant can

then be calculated from

aBG = C\fx - t2 ) /{fa - fa) (11.10)

where

C=l/{(f-Ya )(t-T9 )}. (11.11)

This measurement is repeated 50 to 100 times and an average calculated. Another pair of fas is chosen and

another average determined, eventually covering a 3 dB range on both sides of the optimum /? in steps of

1 dB. An average (aBG) and an estimated standard error of the mean (SEM) for the collection of preceding

averages is then calculated.

The DUT noise temperature is estimated (Tx ) from (11.8) while (11.9) shows that this estimate differs from the "true" DUT temperature (Tx ) by the amount equal to the second term of (11.9). The resulting

error in the DUT temperature is £n =| [fx - Ta )(fx - T.) | en /fx (11.12)

where

e„ =| aBG | +0.64SEM. (11.13)

The second term in (11.13) assures that there is less than a 10% probability that the error is greater than

stated if the distribution of the averages leading to aBG is normal. The current value of en is 1.42 x 10 -8

. A previous determination gave the value of 1.55 x 10 -8

. To monitor the system linearity an attenuator at

the input (the one shown in fig 6) and output (not shown in the figure) of the second i.f. amplifier are switched an equal amount and the output powers checked to insure that the two powers are equal. If the

16
values obtained are within 1 part in 104

, the system is behaving properly. Inserting the value 1.42 x 10 *

into (11.12) yields

£n = 1.42 x 10- 6

| {fx - Ta ){fx - T.) | /tx %. (11.14)

Check standards are also used to monitor the overall system performance.

12. Total Error

The total systematic error is the linear sum of the error components discussed in §3 through §11, and it

is assumed that the true value of the DUT noise temperature lies within the range given by the noise

temperature calculated from the radiometer equation plus or minus this total error (assuming no random

errors).

Et = £» + £a + £y + £m + £at

(12.1)

+ £i + £c + So + £n An unweighted root-sum-square (rss) error Ergs is also defined, where dependent errors are summed linearly

before the rss calculation is performed. Only £y, £m , and £a, are interdependent, so the definition leads to Ertt = [£? + £? + (£y + £m + £as f + £?

(12.2)

+ £l + £l + £l\"\

A typical set of error components is shown in Table 1 for a frequency of 3 GHz and a DUT noise

temperature of 11000 K. The true error is less than the total error (1.72%) at the bottom of the table.

The systematic error for the 1-12 GHz coaxial system at 3 GHz as a function of DUT noise temperature

is shown in figure 7. Figure 8 shows the error as a function of frequency. The thick portions of the linear and

rss curves are a reminder that the measurements leading to these two figures and the table were performed

with the 2-4 GHz front end. Measurements with the other front ends, when available, are expected to yield error limits close to the dashed portions of the two curves in figure 8. The systematic errors for the

millimeterwave systems will be similar to those in the figures.

17
Table 1. A typical set of systematic errors at 3 GHz for a DUT noise temperature of 11000 K. The source

errors in column two are explained in the text. Column three shows the resulting % errors in the

DUT noise temperature.

DUT NOISE TEMPERATURE ERROR

Frequency = 3 GHz Noise Temp = 11000 kelvin

Source

Tm

Ta

Mismatch

Asymmetry

Isolation

Connector

Offset

Nonlinearity

Source Error

eq (3.2)

0.25 K

0.005

0.01 dB

50 dB

0.05/ 1 / 2

eq (10.1)

1.42* 10~ 8

± % Error in Tx

0.64

0.17

0.14

0.22

0.09

0.15

0.07

0.02

Total Error (linear sum) 1.72

RSS Error 0.90

IS. Summary, Discussion, and Conclusions

A noise measurement with the NIST automated radiometer employs the radiometer equation (2.7) to estimate

the noise temperature of a device under test from the calculated ambient and cryogenic noise temperatures,

and the measured Y's, M's, and rj's. The associated errors are summarized in figures 7 and 8 and Table 1, and are discussed in sections 3 thru 12.

Of the few questions still to be answered by future measurements, however, connector error presents the

greatest uncertainty. The 0.05/ 1 ' 2 % error assumed in section 9 for this error is ad hoc at best, and needs more investigation.

The cryogenic standard has been implemented with both a type-N male and an APC 7 mm connector,

so the devices to be tested must have one of these two types of connectors.

Operationally, the radiometer is one of the most stable in existence, and in conjunction with the cryogenic

standard produces very repeatable results [2].

18
14. Acknowledgments

Both the configuration and implementation of the NIST automated radiometer covers a significant time span

and has involved a number of contributors. Among the more important are G. J. Counas, L. D. Driver, C. K. S. Miller, G. R. Reeve, D. F. Wait, and J. Wakefield.

19
ccO

cc

cc

LU

SYSTEMATIC ERROR VS. NOISE TEMPERATURE

(f = 3GHz, Ts = TtK, Ta = 300K, Ill's = 0.1) 2-4 GHz FRONT END

10 100 1000 10000 100000

Tx (kelvins)

Figure 7. Systematic error as a function of DUT noise temperature at 3 GHz assuming 0.1 reflection coeffi- cients for the noise sources and radiometer input ports.

CCO

CC

CCUJ

2.4

2.2

2

1.8

1.6

1.4

1.2

1

0.8

UH£A&--"

SYSTEMATIC ERROR

VS. FREQUENCY

(T2 = 11000K, Ts = 77K, Ta = 300, Ifl's = 0.1)

BSS_—

-

1 8 10 11 12

FREQUENCY (GHz)

Figure 8. Systematic error as a function of frequency for a DUT noise temperature of 11000 K and 0.1

reflection coefficients.

20
15. References

[5

[6

[7

[8a

[8b

[10

[11

[12

Miller, C. K. S.; Daywitt.W. C. The NBS switching radiometers. Nat. Bur. Stand. (U.S.) NBSIR-3004;

1984 May.

Counas, G. J. 2.0 GHz to 4.0 GHz automated radiometer—operation and service manual. Nat. Bur. Stand. (U.S.) NBSIR 83-1697; 1984 January.

IEEE Standard Dictionary of Electrical and Electronic Terms. New York, N. Y.; Wiley-Interscience, a

division of John Wiley and Sons, Inc.;1972.

Miller, C. K. S.; Daywitt, W. C.;Arthur, M. C. Noise standards, measurements, and receiver noise

definitions. Proc. IEEE, Vol. 55, No. 6: 865-877; 1967 June.

Wells, J. S.; Daywitt, W. C.;Miller, C. K. S. Measurement of effective temperatures of microwave noise

sources. IEEE Trans, on I&M, IM-3: 17-18; 1984 March.

Wait, D. F. The sensitivity of the Dicke Radiometer. J. Res. Nat. Bur. Stand. (U.S.) - C. Eng. and

Inst.,Vol. 71C, No. 2,1967 April.

Mumford, W. W.; Scheibe, E. H. Noise Performance Factors in Communications Systems. Dedham,

Massachusetts: Horizon House-Microwave, Inc.; 1962.

Daywitt, W. C. A coaxial noise source for the 1 GHz to 12.4 GHz frequency range. Nat. Bur. Stand.

(U.S.) TN 1074; 1984 March.

Daywitt, W. C. Design and Error Analysis for the WR10 Thermal Noise Standard. Nat. Bur. Stand.

(U.S.) TN 1071; 1983 December.

Wait, D. F. The impact of automation on NBS noise temperature measurements. Proc. IEEE, Vol.74,

No. 1;117-120; 1984 January.

Larsen, N. T. NBS Type-IV power meter, operation and maintenance manual. Nat. Bur. Stand. (U.S.)

NBSIR 77-866;1977 October.

Engen, G. F. A new method of characterizing amplifier noise performance. IEEE Trans, on Instr.,IM-19,

No.4, 344-349; 1970 November.

Southworth, G. C. Principles and Applications of Waveguide Transmission. New York, N. Y.J D. Van

Nostrand Company, Inc.; 1956.

GPO 675-091/5061 21

NBS-114A (REV. 2-8C) U.S. DEPT. OF COMM. BIBLIOGRAPHIC DATA

SHEET (See instructions)

1. PUBLICATION OR

REPORT NO. NIST/TN-1327

2. Performing Organ. Report No 3. Publication Date

March 1989

4. TITLE AND SUBTITLE

Radiometer Equation and Analysis of Systematic Errors for the

NIST Automated Radiometers

5. AUTHOR(S)

W. C. Daywitt

6. PERFORMING ORGANIZATION (If joint or other than NBS, see instructions)

KXXKHHHXBHRXXH)©KXXKMRRRR£ National Institute of Standards

DEPARTMENT OF COMMERCE an(j Technology

WASHINGTON, D.C. 20234

7. Contract/Grant No.

8. Type of Report & Period Covered

9. SPONSORING ORGANIZATION NAME AND COMPLETE ADDRESS (Street. City. State, ZIP)

10. SUPPLEMENTARY NOTES

~2 Document describes a computer program; SF-185, FIPS Software Summary, is attached.

11. ABSTRACT (A 200-word or less (actual summary of most significant information. If document includes a significant

bibliography or literature survey, mention it here)

Equations used in the NIST coaxial and waveguide automated radiometers to

estimate the noise temperature and associated errors of a single-port noise

source are derived in this report. These equations form the foundation upon

which the microwave and millimeterwave noise calibration and special test

services are performed. Results from the 1-12 GHz coaxial radiometer are presented.

12. KEY WORDS (Six to twelve entries; alphabetical order; capitalize only proper names; and separate key words by semicolons)

calibration; error analysis; microwave; millimeterwave; noise temperature;

radiometer

13. AVAILABILITY

[jK)(Un limited Q For Official Distribution. Do Not Release to NTIS

jxx| Order From Superintendent of Documents, U.S. Government Printing Office, Washington, D.C.

20402.

] Order From National Technical Information Service (NTIS), Springfield, VA. 22161

14. NO. OF

PRINTED PAGES

28

15. Price

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