# Patent

## Title
Display device, data driving circuit, and display panel driving method

## Abstract

A display device includes a plurality of selection scan lines, a plurality of current lines, a selection scan driver which sequentially selects the plurality of selection scan lines in each selection period, a data driving circuit which applies a reset voltage to the plurality of current lines in the selection period and supplies a designating current having a current value corresponding to an image signal to the plurality of current lines after applying the reset voltage, and a plurality of pixel circuits which are connected to the plurality of selection scan lines and the plurality of current lines, and supply a driving current having a current value corresponding to the current value of the designating current which flows through the plurality of current lines.

## Background
CROSS-REFERENCE TO RELATED APPLICATIONS
This application is based upon and claims the benefit of priority from prior Japanese Patent Application No. 2004-009146, filed Jan. 16, 2004, the entire contents of which are incorporated herein by reference.
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a display panel driving method of driving a display panel including a light-emitting element for each pixel, a data driving circuit for driving the display panel, and a display device including the display panel, the data driving circuit, and a selection scan driver.
2. Description of the Related Art
Generally, liquid crystal displays are classified into active matrix driving type liquid crystal displays and simple matrix driving type liquid crystal displays. The active matrix driving type liquid crystal displays display images having contrast and resolution higher than those displayed by the simple matrix driving type liquid crystal displays. In the active matrix driving type liquid crystal display, a liquid crystal element which also functions as a capacitor, and a transistor which functions as a pixel switching element are formed for each pixel. In the active matrix driving system, when a voltage at a level representing luminance is applied to a current line by a data driver while a scan line is selected by a scan driver serving as a shift register, this voltage is applied to the liquid crystal element via the transistor. Even when the transistor is turned off in a period after the selection of the scan line is complete and before the scan line is selected again, the liquid crystal element functions as a capacitor, so the voltage level is held in this period. As described above, the light transmittance of the liquid crystal element is refreshed while the scan line is selected, and light from a backlight is transmitted through the liquid crystal element having the refreshed light transmittance. In this manner, the liquid crystal display expresses a tone.
Displays using organic EL (ElecctroLuminescent) elements as self-light-emitting elements require no such a backlight as used in the liquid crystal displays, and hence are optimum for flat display devices. In addition, the viewing angle is not limited unlike in the liquid crystal display. Therefore, these organic EL displays are increasingly expected to be put into practical use as next-generation display devices.
From the viewpoints of high luminance, high contrast, and high resolution, active matrix driving type organic EL displays are developed similarly to the liquid crystal displays. For example, in the conventional active matrix driving type organic EL display described in Jpn. Pat. Appln. KOKAI Publication No. 2000-221942, a pixel circuit (referred to as an organic EL element driving circuit in patent reference 1) is formed for each pixel. This pixel circuit includes an organic EL element, driving TFT, first switching element, switching TFT, and the like. When a control line is selected, a current source driver applies a voltage as luminance data to the gate of the driving TFT. Consequently, the driving TFT is turned on, and a driving current having a current value corresponding to the level of the gate voltage flows from a power supply line to the driving TFT via the organic EL element, so the organic EL element emits light at luminance corresponding to the current value of the electric current. When the selection of the control line is complete, the gate voltage of the driving TFT is held by the first switching element, so the emission of the organic EL element is also held. When a blanking signal is input to the gate of the switching TFT after that, the gate voltage of the driving TFT decreases to turn it off, and the organic EL element is also turned off to complete one frame period.
Generally, the channel resistance of a transistor changes in accordance with a change in ambient temperature, or changes when the transistor is used for a long time. As a consequence, the gate threshold voltage changes with time, or differs from one transistor to another. Therefore, in the conventional voltage-controlled, active matrix driving type organic EL display in which the luminance and tone are controlled by the signal voltage, it is difficult to uniquely designate the current value of an electric current which flows through the organic EL element by the level of the gate voltage of the driving TFT, even if the current value of the electric current which flows through the organic EL element is changed by changing the level of the gate voltage of the driving TFT by using the signal voltage from the current line. That is, even when the gate voltage having the same level is applied to the driving TFTs of a plurality of pixels, the luminance of the organic EL element changes from one pixel to another. This produces variations in luminance on the display screen. Also, since the driving TFT deteriorates with time, the same gate voltage as the initial gate voltage cannot generate a driving current having the same current value as the initial current value. This also varies the luminance of the organic EL elements.
BRIEF SUMMARY OF THE INVENTION
It is, therefore, an object of the present invention to provide a display device, data driving circuit, and display panel driving method capable of displaying high-quality images.
A display device according to an aspect of the present invention comprises, a plurality of selection scan lines;
a plurality of current lines;
a selection scan driver which sequentially selects the plurality of selection scan lines in each selection period;
a data driving circuit which applies a reset voltage to the plurality of current lines in a first part of the selection period, and supplies a designating current having a current value corresponding to an image signal to the plurality of current lines in a second part of the selection period after applying the reset voltage in the selection period; and
a plurality of pixel circuits which are connected to the plurality of selection scan lines and the plurality of current lines, and supply a driving current having a current value corresponding to the current value of the designating current which flows through the plurality of current lines.
A display device according to another aspect of the present invention comprises, a plurality of selection scan lines;
a plurality of current lines;
a plurality of light-emitting elements which are arranged at intersections of the plurality of selection scan lines and the plurality of current lines, and emit light at luminance corresponding to a current value of a driving current;
a selection scan driver which sequentially select the plurality of selection scan lines in each selection period;
a data driving circuit which applies a reset voltage to the plurality of current lines in a first part of the selection period, and supplies a designating current having a current value corresponding to an image signal to the plurality of current lines in a second part of the selection period after applying the reset voltage in the selection period; and
a plurality of pixel circuits which are connected to the plurality of selection scan lines and the plurality of current lines, and electrically connect the plurality of current lines and the plurality of light-emitting elements to each other in the selection period.
A data driving circuit according to still another aspect of the present invention comprises, a plurality of light-emitting elements connected to a plurality of selection scan lines and a plurality of current lines, a selection scan driver which sequentially selects the plurality of selection scan lines in each selection period, and a plurality of pixel circuits connected to the plurality of light-emitting elements,
wherein a reset voltage is applied to the plurality of current lines in a first part of the selection period, and a designating current having a current value corresponding to an image signal is supplied to the plurality of current lines in a second part of the selection period after the first part of the selection period.
A display panel driving method according to still another aspect of the present invention comprises, a selection step of sequentially selecting a plurality of selection scan lines of a display panel comprising a plurality of pixel circuits connected to the plurality of selection scan lines and a plurality of current lines, and a plurality of light-emitting elements which are arranged at intersections of the plurality of selection scan lines and the plurality of current lines, each of the light-emitting elements emits light at luminance corresponding to a current value of a current flowing the current line; and
a reset step of applying a reset voltage to the plurality of current lines in an initial part of a period in which each of the plurality of selection scan lines is selected.
In the present invention, it is possible not only to discharge the parasitic capacitance of a current line by applying a reset voltage in a selection period, but also to discharge the parasitic capacitance of a pixel circuit or the parasitic capacitance of a light-emitting element.



BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING
FIG. 1 is a block diagram of an organic electroluminescent display 1 according to the first embodiment of the present invention;
FIG. 2 is a plan view of a pixel Pi,j of the organic electroluminescent display 1;
FIG. 3 is an equivalent circuit diagram of four adjacent pixels Pi,j, Pi+1,j, Pi,j+1, and Pi+1,j+1 of the organic electroluminescent display 1;
FIG. 4 is a timing chart showing the levels of signals in the organic electroluminescent display 1;
FIG. 5 is a graph showing the current-voltage characteristics of an N-channel field-effect transistor;
FIG. 6 shows an equivalent circuit diagram of two adjacent pixels Pi,j and Pi,j+1 in the ith row, and the states of electric currents and voltages in a reset period TR of the ith row;
FIG. 7 shows the equivalent circuit diagram of the two adjacent pixels Pi,j and Pi,j+1 in the ith row, and the states of electric currents and voltages after the reset period TR in a selection period TSE of the ith row;
FIG. 8 shows the equivalent circuit diagrams of the two adjacent pixels Pi,j and Pi,j+1 in the ith row, and the states of electric currents and voltages in a non-selection period TNSE of the ith row;
FIG. 9 is a timing chart showing the levels of electric currents and voltages pertaining to the pixel Pi,j;
FIG. 10 is a block diagram of an organic electroluminescent display according to the second embodiment of the present invention;
FIG. 11 is a block diagram of an organic electroluminescent display according to the third embodiment of the present invention; and
FIG. 12 is a block diagram of an organic electroluminescent display according to the fourth embodiment of the present invention.



DETAILED DESCRIPTION OF THE INVENTION
Best modes for carrying out the invention will be described below with reference to the accompanying drawings. Various technically preferred limitations are imposed on the following embodiments in order to, carry out the present invention. However, the scope of the invention is not limited to the embodiments and examples shown in the drawing.
First Embodiment
FIG. 1 is a block diagram showing an organic electroluminescent display 1 according to the first embodiment to which the organic electroluminescent display of the present invention is applied. As shown in FIG. 1, the organic electroluminescent display 1 includes, as its basic configuration, an organic electroluminescent display panel 2 having m selection scan lines X1 to Xm, m voltage supply lines Z1 to Zm, n current lines Y1 to Yn, and pixels P1,1 to Pm,n. The display 1 further includes, a scan driving circuit 9 for linearly scanning the organic electroluminescent display panel 2 in the longitudinal direction, and a data driving circuit 7 for supplying a tone designating current IDATA to the current lines Y1 to Yn in cooperation with the scan driving circuit 9. Here, each of m and n is a natural number of 2 or more.
The scan driving circuit 9 has a selection scan driver 5 for sequentially selecting the selection scan lines X1 to Xm, and a voltage supply driver 6 for sequentially selecting the voltage supply lines Z1 to Zm in synchronism with the sequential selection of the selection scan lines X1 to Xm by the selection scan driver 5. The data driving circuit 7 has a current source driver 3. The driver 3 includes n current terminals CT1 to CTn and allows the tone designating current IDATA to flow through the current terminals CT1 to CTn, and switches S1 to Sninterposed between the current terminals CT1 to CTn and current lines Y1 to Yn.
The organic electroluminescent display panel 2 has a structure in which a display unit 4 for practically displaying images is formed on a transparent substrate. The selection scan driver 5, voltage supply driver 6, current source driver 3, and switches S1 to Sn are arranged around the display unit 4. Portions or the whole of the selection scan driver 5, the voltage supply driver 6, the current source driver 3, and at least one of the switches S1 to Sn can be integrated with the organic electroluminescent display panel 2 as they are formed on the transparent substrate, or can be formed around the organic electroluminescent display panel 2 as they are formed into a chip different from the organic electroluminescent display panel 2. Note that the display unit 4 may also be formed on a flexible sheet such as a resin sheet, instead of the transparent substrate.
In the display unit 4, the (m×n) pixels P1,1 to Pm,n are formed in a matrix on the transparent substrate such that m pixels are arranged in the longitudinal direction, i.e., the column direction, and n pixels are arranged in the lateral direction, i.e., the row direction. A pixel which is an ith pixel (i.e., a pixel in the ith row) from above and a jth pixel (i.e., a pixel in the jth column) from left is a pixel Pi,j. Note that i is a given natural number from 1 to m, and j is a given natural number from 1 to n.
Accordingly, in the display unit 4, the m selection scan lines X1 to Xm running in the row direction are formed parallel to each other on the transparent substrate. The m voltage supply lines Z1 to Zm running in the row direction are formed parallel to each other on the transparent substrate in one-to-one correspondence with the selection scan lines X1 to Xm. The voltage supply line Zk (1≦k≦m−1) is positioned between the selection scan lines Xk and Xk+1, and the selection scan line Xm is positioned between the voltage supply lines Zm−1 and Zm. Also, the n current lines Y1 to Yn running in the column direction are formed parallel to each other on the upper side of the transparent substrate. The selection scan lines X1 to Xm, voltage supply lines Z1 to Zm, and current lines Y1 to Yn are insulated from each other as they are separated by insulating films or the like interposed between them. The n pixels Pi,1 to Pi,n arranged along the row direction are connected to the selection scan line Xi and voltage supply line Zi in the ith row. The m pixels P1,j to Pm,j arranged along the column direction are connected to the current line Yj in the jth column. The pixel Pi,j is positioned at the intersection of the selection scan line Xi and current line Yj. The selection scan lines X1 to Xm are connected to output terminals of the selection scan driver 5. The voltage supply lines Z1 to Zm are connected to output terminals of the voltage supply driver 6.
The pixels P1,1 to Pm,n will be explained below with reference to FIGS. 2 and 3. FIG. 2 is a plan view showing the pixel Pi,j. FIG. 3 is an equivalent circuit diagram showing, e.g., four adjacent pixels Pi,j, Pi+1,j, Pi,j+1, and Pi+1,j+1. FIG. 2 principally shows the electrodes in the pixel Pi,j to allow better understanding.
The pixel Pi,j includes an organic electroluminescent element Ei,j as a self-light-emitting element which emits light in accordance with the value of an electric current, and a pixel circuit Di,j which is formed around the organic electroluminescent element Ei,j, and drives it. Note that the organic electroluminescent element will be referred to as an organic EL element hereinafter.
The organic EL element Ei,j has a stacked structure in which a pixel electrode 51, organic EL layer 52, and common electrode are stacked in this order on the transparent substrate. The pixel electrode 51 functions as an anode. The organic EL layer 52 functions as a light-emitting layer in a broad sense, i.e., transports holes and electrons injected by an electric field, recombines the transported holes and electrons, and emits light by excitons produced by the recombination. The common electrode functions as a cathode. Although the common electrode is formed to cover the entire pixel, the it is not shown in FIG. 2 so that the pixel electrode 51, organic EL layer 52, pixel circuit Di,j and the like are readily seen.
The pixel electrode 51 is patterned for each of the pixels P1,1 to Pm,n in each of regions surrounded by the current lines Y1 to Yn, selection scan lines X1 to Xm, and voltage supply lines Z1 to Zm.
The pixel electrode 51 is a transparent electrode. That is, the pixel electrode 51 has both conductivity and transparency to visible light. Also, the pixel electrode 51 preferably has a relatively high work function, and efficiently injects holes into the organic EL layer 52. Examples of main components of the pixel electrode 51 are tin-doped indium oxide (ITO), zinc-doped indium oxide, indium oxide (In2O3), tin oxide (SnO2), zinc oxide (ZnO), and cadmium-tin oxide (CTO).
The organic EL layer 52 is formed on each pixel electrode 51. The organic EL layer 52 is also patterned for each of the pixels P1,1 to Pm,n. The organic EL layer 52 contains a light-emitting material (phosphor) as an organic compound. This light-emitting material can be either a high- or low-molecular material. In particular, the organic EL layer 52 has a two-layered structure in which a hole transporting layer and a light-emitting layer in a narrow sense are stacked in this order on the pixel electrode 51. The hole transporting layer is made of a PEDOT (polythiophene) as a conductive polymer, and PSS (polystyrene sulfonic acid) as a dopant. The light-emitting layer in a narrow sense is made of a polyfluorene-based, light-emitting material. Note that the organic EL layer 52 may also have a three-layered structure having a hole transporting layer, a light-emitting layer in a narrow sense, and an electron transporting layer stacked in this order on the pixel electrode 51, or a single-layered structure having only a light-emitting layer in a narrow sense, instead of the two-layered structure. An electron or hole injecting layer may also be interposed between appropriate layers in any of these layered structures, and some other stacked structure may also be used.
The organic EL display panel 2 can display full-color images or multicolor images. The organic EL layer 52 of each of the pixels P1,1 to Pm,n is a light-emitting layer in a broad sense which has a function of emitting red, green, or blue light. That is, the organic EL layers 52 which emit red light, green light, and blue light are regularly arranged, and the display unit 4 displays images in a color tone obtained by properly synthesizing these colors.
The organic EL layer 52 is desirably made of an organic compound which is neutral with respect of electrons. This allows balanced injection and transportation of holes and electrons in the organic EL layer 52. One or both of an electron transporting substance and hole transporting substance may also be properly mixed in the light-emitting layer in a narrow sense. It is also possible to cause a charge transporting layer which is an electron or hole transporting layer to function as a recombination region which recombines electrons and holes, and to emit light by mixing a phosphor in this charge transporting layer.
The common electrode formed on the organic EL layers 52 is formed for all the pixels P1,1 to Pm,n. Note that instead of this common electrode formed for all the pixels P1,1 to Pm,n, it is also possible to use a plurality of divided electrodes, e.g., a plurality of stripe electrodes divided for individual columns, or a plurality of stripe electrodes divided for individual rows. Generally, the organic EL layers 52 which emit different colors are made of different materials, and the light emission characteristics with respect to the current density depend upon the material. To adjust the luminance balance between different emission colors, therefore, pixels which emit the same color can be connected together in order to set the value of an electric current for each emission color of the organic EL layer 52. That is, assuming that a first-emission-color pixel emits a predetermined luminance at a relatively low current density, and a second-emission-color pixel requires a high current density in order to emit the same luminance as the first-emission-color pixel, the emission color balance can be adjusted by supplying, to the second-emission-color pixel, a tone electric current which is larger than that of the first-emission-color pixel.
The common electrode is electrically insulated from the selection scan lines X1 to Xm, current lines Y1 to Yn, and voltage supply lines Z1 to Zm. The common electrode is made of a material having a low work function. For example, the common electrode is made of indium, magnesium, calcium, lithium, barium, a rare earth metal, or an alloy containing at least one of these elements. Also, the common electrode can have a stacked structure in which layers of the various materials described above are stacked, or a stacked structure in which a metal layer is deposited in addition to these layers of the various materials. Practical examples are a stacked structure including a low-work-function, high-purity barium layer formed in the interface in contact with the organic EL layer 52, and an aluminum layer which covers this barium layer, and a stacked structure having a lithium layer as a lower layer and an aluminum layer as an upper layer. When the pixel electrode 51 is a transparent electrode and light emitted from the organic EL layer 52 is output from the transparent substrate through the pixel electrode 51, the common electrode preferably has light-shielding properties with respect to the light emitted from the organic EL layer 52, and more preferably has a high reflectance to the light emitted from the organic EL layer 52.
When a forward bias voltage (by which the voltage of the pixel electrode 51 becomes higher than that of the common electrode) is applied between the pixel electrode 51 and common electrode in the organic EL element Ei,j having the stacked structure as described above, holes are injected into the organic EL layer 52 from the pixel electrode 51, and electrons are injected into the organic EL layer 52 from the common electrode. The organic EL layer 52 transports these holes and electrons, and recombines them to produce excitons. Since these excitons excite the organic EL layer 52, the organic EL layer 52 emits light.
The luminance of the organic EL element Ei,j depends on the current value of an electric current which flows through the organic EL element Ei,j; the larger the electric current which flows through the organic EL element Ei,j, the higher the luminance of the organic EL element Ei,j. That is, if deterioration of the organic EL element Ei,j is not taken into consideration, the luminance of the organic EL element Ei,j is uniquely determined when the current value of the electric current which flows through the organic EL element Ei,j is determined.
Each of the pixel circuits D1,1 to Dm,n includes three thin-film transistors (to be simply referred to as transistors hereinafter) 21, 22, and 23, and a capacitor 24.
Each of the transistors 21, 22, and 23 is an N-channel MOS field-effect transistor having a gate, drain, source, semiconductor layer 44, impurity-dosed semiconductor layer, and gate insulating film. Each transistor is particularly an a-Si transistor in which the semiconductor layer 44 (channel region) is made of amorphous silicon. However, each transistor may also be a p-Si transistor in which the semiconductor layer 44 is made of polysilicon. In either case, the transistors 21, 22, and 23 are N-channel field-effect transistors, and can have either an inverted stagger structure or a coplanar structure.
Also, the transistors 21, 22, and 23 can be simultaneously formed in the same process. In this case, the compositions of the gates, drains, sources, semiconductor layers 44, impurity-dosed semiconductor layers, and gate insulating films of the transistors 21, 22, and 23 are the same, and the shapes, sizes, dimensions, channel widths, and channel lengths of the transistors 21, 22, and 23 are different from each other in accordance with the functions of the transistors 21, 22, and 23. Note that the transistors 21, 22, and 23 will be referred to as a first transistor 21, second transistor 22, and driving transistor 23, respectively, hereinafter.
The capacitor 24 has a first electrode 24A connected to a gate 23g of the driving transistor 23, a second electrode 24B connected to a source 23s of the transistor 23, and a gate insulating film (dielectric film) interposed between these two electrodes. The capacitor 24 has a function of storing electric charges between the gate 23g and source 23s of the driving transistor 23.
In the second transistor 22 of each of the pixel circuits Di,1 to Di,n in the ith row, a gate 22g is connected to the selection scan line Xi in the ith row, and a drain 22d is connected to the voltage supply line Zi in the ith row. In the driving transistor 23 of each of the pixel circuits Di,1 to Di,n in the ith row, a drain 23d is connected to the voltage supply line Zi in the ith row through a contact hole 26. In the first transistor 21 of each of the pixel circuits Di,1 to Di,n in the ith row, a gate 21g is connected to the selection scan line Xi in the ith row. In the first transistor 21 of each of the pixel circuits D1,j to Dm,j in the jth column, a source 21s is connected to the current line Yj in the jth column.
In each of the pixels P1,1 to Pm,n, a source 22s of the second transistor 22 is connected to the gate 23g of the driving transistor 23 through a contact hole 25, and to one electrode of the capacitor 24. The source 23s of the driving transistor 23 is connected to the other electrode of the capacitor 24, and to a drain 21d of the first transistor 21. The source 23s of the driving transistor 23, the other electrode of the capacitor 24, and the drain 21d of the first transistor 21 are connected to the pixel electrode 51.
The voltage of the common electrode of the organic EL elements E1,1, to Em,n is held at a predetermined reference voltage VSS. In this embodiment, the reference voltage VSS is set at 0 [V] by grounding the common electrode of the organic EL elements E1,1, to Em,n.
The pixel electrodes 51 are divided by patterning for individual pixels surrounded by regions surrounded by the current lines Y1 to Yn, selection scan lines X1 to Xm, and voltage supply lines Z1 to Zm. In addition, the edges of each pixel electrode 51 are covered with an interlayer dielectric film made of silicon nitride or silicon oxide which covers the three transistors 21, 22, and 23 of each pixel circuit, and the upper surface of the center of the pixel electrode 51 is exposed through a contact hole 55 formed in this interlayer dielectric film. Note that the interlayer dielectric film can have a first layer made of silicon nitride or silicon oxide, and a second layer formed on the first layer by using an insulating film made of, e.g., polyimide.
Between the selection scan line Xi and current line Yj, and between the voltage supply line Zi and current line Yj, a protective film 44A is formed by patterning the same film as the semiconductor layer 44 of each of the transistors 21 to 23, in addition to the gate insulating film. Note that in order to protect the surface, which serves as a channel, of the semiconductor layer 44 of each of the transistors 21, 22, and 23 from being roughened by an etchant used in patterning, a blocking insulating layer made of silicon nitride or the like may also be formed except for the two end portions of the semiconductor layer 44. In this case, a protective film may be formed by patterning the same film as the blocking insulating layer between the selection scan line Xi and current line Yj, and between the voltage supply line Zi and current line Yj. This protective film and the protective film 44A may also be overlapped.
The selection scan driver 5, voltage supply driver 6, switches S1 to Sn, and current source driver 3 will be described below with reference to FIG. 4. FIG. 4 is a timing chart showing, from above, the voltage of the selection scan line X1, the voltage of the voltage supply line Z1, the voltage of the selection scan line X2, the voltage of the voltage supply line Z2, the voltage of the selection scan line X3, the voltage of the voltage supply line Z3, the voltage of the selection scan line Xm, the voltage of the voltage supply line Zm, the level (voltage value) of a switching signal inv.Φ, the level of a switching signal Φ, the voltage of the current line Yj, the voltage of the pixel electrode 51 of the organic EL element E1,j, the luminance of the organic EL element E1,j, the voltage of the pixel electrode 51 of the organic EL element E2,j, and the luminance of the organic EL element E2,j. Referring to FIG. 4, the abscissa represents the common time.
The selection scan driver 5 is a so-called shift register, and has an arrangement in which m flip-flop circuits and the like are connected in series. That is, the selection scan driver 5 sequentially selects the selection scan lines X1 to Xm by sequentially outputting selection signals in order from the selection scan line X1 to the selection scan line Xm (the selection scan line Xm is followed by the selection scan line X1), thereby sequentially selecting the first and second transistors 21 and 22 in these rows connected to the selection scan lines X1 to Xm.
More specifically, as shown in FIG. 4, the selection scan driver 5 individually applies, to the selection scan lines X1 to Xm, a high-level (ON-level) ON voltage VON (much higher than the reference voltage VSS) as a selection signal or a low-level OFF voltage VOFF (equal to or lower than the reference voltage VSS) as a non-selection signal, thereby sequentially selecting the selection scan lines X1 to Xm.
That is, when the selection scan driver 5 applies the ON voltage VON to the selection scan line Xi, the selection scan line Xi in the ith row is selected. A period in which the selection scan driver 5 applies the ON voltage VON to the selection scan line Xi in the ith row and thereby selects the selection scan line Xi in the ith row is called a selection period TSE of the ith row. Note that while applying the ON voltage VON to the selection scan line Xi, the selection scan driver 5 applies the OFF voltage VOFF to the other selection scan lines X1 to Xm (except for the selection scan line Xi). Accordingly, the selection periods TSE of the selection scan lines X1 to Xm do not overlap each other.
When the selection scan driver 5 applies the ON voltage VON to the selection scan line Xi in the ith row, the first and second transistors 21 and 22 are turned on in each of the pixel circuits Di,1 to Di,n connected to the selection scan line Xi in the ith row. Since the first transistors 21 are turned on, an electric current which flows through the current lines Y1 to Yn can flow through the pixel circuits Di,1 to Di,n.
After the selection period TSE in which the selection scan line Xi in the ith row is selected, the selection scan driver 5 applies the OFF voltage VOFF to the selection scan line Xi to cancel the selection of the selection scan line Xi. As a consequence, in each of the pixel circuits Di,1 to Di,n connected to the selection scan line Xi in the ith row, the first and second transistors 21 and 22 are turned off. Since the first transistors 21 are turned off, the electric current which flows through the current lines Y1 to Yn cannot flow through the pixel circuits Di,1 to Di,n any longer. Note that a period in which the selection scan driver 5 applies the OFF voltage VOFF to the selection scan line Xi in the ith row and thereby keeps the selection scan line Xi in the ith row unselected is called a non-selection period TNSE of the ith row. In this case, a period represented by TSE+TNSE=TSC, i.e., a period from the start time of the selection period TSE of the selection scan line Xi in the ith row to the start time of the next selection period TSE of the selection scan line Xi in the ith row, is one frame period of the ith row.
The voltage supply driver 6 is a so-called shift register, and has an arrangement in which m flip-flop circuits are connected in series. That is, in synchronism with the selection scan driver 5, the voltage supply driver 6 sequentially selects the voltage supply lines Z1 to Zm by sequentially outputting selection signals in order from the voltage supply line Z1 to the voltage supply line Zm (the voltage supply line Zm is followed by the voltage supply line Z1), thereby sequentially selecting the driving transistors 23 in these rows connected to the voltage supply lines Z1 to Zm.
More specifically, as shown in FIG. 4, the voltage supply driver 6 individually supplies, to the voltage supply lines Z1 to Zm, a low-level tone designating current reference voltage VLOW (which is equal to or lower than the reference voltage VSS) as a selection signal or a high-level driving current reference voltage VHIGH (which is higher than both the reference voltage VSS and tone designating current reference voltage VLOW) as a non-selection signal, thereby sequentially selecting the voltage supply lines Z1 to Zm.
That is, in the selection period TSE in which the selection scan line Xi in the ith row is selected, the voltage supply driver 6 applies the tone designating current reference voltage VLOW to the voltage supply line Zi in the ith row, thereby selecting the voltage supply line Zi in the ith row. While applying the tone designating current reference voltage VLOW to the voltage supply line Zi, the voltage supply driver 6 applies the driving current reference voltage VHIGH to the other voltage supply lines Z1 to Zm (except for the voltage supply line Zi).
On the other hand, in the non-selection period TNSE in which the selection scan line Xi in the ith row is not selected, the voltage supply driver 6 applies the driving current reference voltage VHIGH to the voltage supply line Zi to cancel the selection of the voltage supply line Zi in the ith row. Since the driving current reference voltage VHIGH is higher than the reference voltage VSS, an electric current flows from the voltage supply line Zi to the organic EL element Ei,j if the driving transistor 23 is ON and the transistor 21 is OFF.
The tone designating current reference voltage VLOW applied by the voltage supply driver 6 is equal to or lower than the reference voltage VSS. Therefore, even when the driving transistor 23 of each of the pixels P1,1 to Pm,n is turned on in the selection period TSE, a zero voltage or reverse bias voltage is applied between the anode and cathode of each of the organic EL elements E1,1 to Em,n. Accordingly, no electric current flows through the organic EL elements E1,1 to Em,n in the selection period TSE, so the organic EL elements E1,1 to Em,n do not emit light. On the other hand, the driving current reference voltage VHIGH applied by the voltage supply driver 6 is higher than the reference voltage VSS. As shown in FIG. 5, the driving current reference voltage VHIGH is so set that a source-to-drain voltage VDS of the driving transistor 23 is in a saturated region. Accordingly, when the driving transistors 23 are ON in the non-selection period TNSE, a forward bias voltage is applied to the organic EL elements E1,1, to Em,n. In the non-selection period TNSE, therefore, an electric current flows through the organic EL elements E1,1 to Em,n, and the organic EL elements E1,1 to Em,n emit light.
The driving current reference voltage VHIGH will be explained below. FIG. 5 is a graph showing the current-voltage characteristics of the N-channel field-effect transistor. Referring to FIG. 5, the abscissa indicates the divided voltage of the driving transistor and the divided voltage of the organic EL element connected in series to the driving transistor, and the ordinate indicates the current value of an electric current in the drain-to-source path. In an unsaturated region (a region where source-to-drain voltage VDS<drain saturated threshold voltage VTH: the drain saturated threshold voltage VTH is a function of a gate-to-source voltage VGS, and is uniquely determined by the gate-to-source voltage VGS if the gate-to-source voltage VGS is determined) shown in FIG. 5, if the gate-to-source voltage VGS is constant, a drain-to-source current IDS increases as the source-to-drain voltage VDS increases. In addition, in a saturated region (in which source-to-drain voltage VDS≧drain saturated threshold voltage VTH) shown in FIG. 5, if the gate-to-source voltage VGS is constant, the drain-to-source current IDS is substantially constant even when the source-to-drain voltage VDS increases.
Also, in FIG. 5, gate-to-source voltages VGS1 to VGSMAX have the relationship 0 [V]<VGS1<VGS2<VGS3<VGS4<VGSMAX. That is, as is apparent from FIG. 5, if the source-to-drain voltage VDS is constant, the drain-to-source current IDS increases in both the unsaturated and saturated regions as the gate-to-source voltage VGS increases. In addition, the drain saturated threshold voltage VTH increases as the gate-to-source voltage VGS increases.
From the foregoing, in the unsaturated region, the drain-to-source current IDS changes if the source-to-drain voltage VDS slightly changes while the gate-to-source voltage VGS is constant. In the saturated region, however, the drain-to-source current IDS is uniquely determined by the gate-to-source voltage VGS.
The drain-to-source current IDS when the maximum gate-to-source voltage VGSMAX is applied to the driving transistor 23 is set to be an electric current which flows between the common electrode and the pixel electrode 51 of the organic EL element Ei,j which emits light at the maximum luminance.
Also, the following equation is met so that the driving transistor 23 maintains the saturated region in the selection period TSE even when the gate-to-source voltage VGS of the driving transistor 23 is the maximum voltage VGSMAX in the non-selection period.

VLOW=VHIGH−VE−VSS≧VTHMAX 

where VE is the anode-to-cathode voltage which the organic EL element Ei,j requires to emit light at the maximum luminance in the light emission life period, and VTHMAX is the source-to-drain saturated voltage level of the driving transistor 23 when the voltage is VGSMAX. The driving current reference voltage VHIGH is set to satisfy the above equation. Accordingly, even when the source-to-drain voltage VDS of the driving transistor 23 decreases by the divided voltage of the organic EL element Ei,j connected in series to the driving transistor 23, the source-to-drain voltage VDS always falls within the range of the saturated state, so the drain-to-source current IDS is uniquely determined by the gate-to-source voltage VGS.

As shown in FIGS. 1 and 3, the current lines Y1 to Yn are connected to the current terminals CT1 to CTn of the current source driver 3 via the switches S1 to Sn. An 8-bit digital tone image signal is input to the current source driver 3. This digital tone image signal input to the current source driver 3 is converted into an analog signal by an internal D/A converter of the current source driver 3. The current source driver 3 generates, at the current terminals CT1 to CTn, a tone designating current IDATA having a current value corresponding to the converted analog signal. As shown in FIG. 4, the current source driver 3 controls the current value of the tone designating current IDATA at the current terminals CT1 to CTn in accordance with the image signal for each selection period TSE of each row, and holds the current value of the tone designating current IDATA constant in a period from the end of each reset period TR to the end of the corresponding selection period TSE. The current source driver 3 supplies the tone designating current IDATA from the current lines Y1 to Yn to the current terminals CT1 to CTn via the switches S1 to Sn. As shown in FIGS. 1 and 3, the switches S1 to Sn are connected to the current lines Y1 to Yn, and the current terminals CT1 to CTn of the current source driver 3 are connected to the switches S1 to Sn. In addition, the switches S1 to Sn are connected to a reset input terminal 41, and a reset voltage VR is applied to the switches S1 to Sn via the reset input terminal 41. The switches S1 to Sn are also connected to a switching signal input terminal 42, and a switching signal Φ is input to the switches S1 to Sn via the switching signal input terminal 42. Furthermore, the switches S1 to Sn are connected to a switching signal input terminal 43, and a switching signal inv.Φ obtained by inverting the switching signal Φ is input to the switches S1 to Sn via the switching signal input terminal 43. The reset voltage VR is constant and has the same level (voltage value) as the tone designating current reference voltage VLOW. More specifically, the reset voltage VR is set at 0 [V] by grounding the reset input terminal 41. Especially when the reset voltage VR of the ith row is made equal to the voltage of the voltage supply line Zi in the ith row in the selection period TSE, the voltages of the electrodes 24A and 24B of the capacitor 24 become equal to each other. Consequently, the capacitor 24 is discharged, so the gate-to-source voltage of the driving transistor 23 is set at 0V.
The switch Sj (which is interposed between the current line Yj in the jth column and the current terminal CTj in the jth column) switches the state in which the current source driver 3 supplies the tone designating current IDATA to the current line Yj, and the state in which the reset voltage VR is applied to the current line Yj. That is, as shown in FIG. 4, if the switching signal Φ is at high level and the switching signal inv.Φ is at low level, the switch Sj shuts off the electric current of the current terminal CTj, and applies the reset voltage VR to the current line Yj, the drain 21d of the first transistor 21, the electrode 24B of the capacitor 24, the source 23s of the driving transistor 23, and the pixel electrode 51 of the organic EL element Ex,j (1≦x≦m), thereby discharging the electric charge stored in these components in the preceding selection period TSE. On the other hand, if the switching signal Φ is at low level and the switching signal inv.Φ is at high level, the switch Sj allows the electric current of the current terminal CTj to flow through the current line Yj, and shuts down the application of the reset voltage VR to the current line Yj.
The cycle of the switching signals Φ and inv.Φ will be explained below. As shown in FIG. 4, the cycle of the switching signals Φ and inv.Φ is the same as the selection period TSE. That is, when the selection scan driver 5 starts applying the ON voltage VON to one of the selection scan lines X1 to Xm (i.e., when the selection period TSE of each row starts), the switching signal Φ changes from high level to low level, and the switching signal inv.Φ changes from low level to high level. While the selection scan driver 5 is applying the ON voltage VON to one of the selection scan lines X1 to Xm (i.e., in the selection period TSE of each row), the switching signal Φ changes from low level to high level, and the switching signal inv.Φ changes from high level to low level. A period in which the switching signal Φ is at high level and the switching signal inv.Φ is at low level in the selection period TSE of the selection scan line Xi in the ith row is called the reset period TR of the ith row.
An example of the switch Sj will be explained below. The switch Sj is made up of first and second N-channel field-effect transistors 31 and 32. The gate of the first transistor 31 is connected to the switching signal input terminal 43, and thus the switching signal inv.Φ is input to the gate of the transistor 31. Also, the gate of the second transistor 32 is connected to the switching signal input terminal 42, and thus the switching signal Φ is input to the gate of the transistor 32. The drain of the first transistor 31 is connected to the current line Yj, and the source of the transistor 31 is connected to the current terminal CTj. The drain of the transistor 32 is connected to the current line Yj. The source of the transistor 32 is connected to the reset input terminal 41, and the reset voltage VR which is a constant voltage is applied to the source of the transistor 32. In this arrangement, when the switching signal Φ is at high level and the switching signal inv.Φ is at low level, the transistor 32 is turned on, and the transistor 31 is turned off. When the switching signal Φ is at low level and the switching signal inv.Φ is at high level, the transistor 31 is turned on, and the transistor 32 is turned off. The transistors 31 and 32 can be fabricated in the same steps as the transistors 21 to 23 of the pixel circuits D1,1 to Dm,n.
The functions of the pixel circuits D1,1 to Dm,n will be described below with reference to FIGS. 6 to 8. In FIGS. 6 to 8, the flows of electric currents are indicated by arrows.
FIG. 6 is a circuit diagram showing the states of the voltages in the reset period TR of the selection period TSE of the ith row. As shown in FIG. 6, in the reset period TR of the ith row, the selection scan driver 5 applies the ON voltage VON to the selection scan line Xi, and the voltage supply driver 6 applies the tone designating current reference voltage VLOW to the voltage supply line Zi. In addition, in the reset period TR of the ith row, the switches S1 to Sn apply the reset voltage VR to the current lines Y1 to Yn. In the reset period TR of the ith row, therefore, the first transistors 21 of the pixel circuits Di,1 to Di,n are ON. Consequently, as shown in FIG. 4, the voltages of the pixel electrodes 51 of the organic EL elements Ei,1 to Ei,n, the drains 21d of the first transistors 21 in the ith row, the electrodes 24B of the capacitors 24 in the ith row, the sources 23s of the driving transistors 23 in the ith row, and the current lines Y1 to Yn are set in a steady state by the reset voltage VR, thereby discharging the electric charge stored by these parasitic capacitances in the preceding selection period TSE. Accordingly, the tone designating current IDATA having a steady current value can be rapidly written in the next selection period TSE.
The parasitic capacitances of the organic. EL elements Ei,1 to Ei,n are particularly large. Therefore, when the tone designating current IDATA having a low current value is written, it takes a long time to make the current value steady by resetting the electric charge written in the organic EL element in the preceding frame period TSC if the reset voltage VR is not applied in the selection period TSE. However, the reset voltage VR is forcedly applied in the selection period TSE, so the parasitic capacitance of the organic EL element can be rapidly discharged. Also, when the reset voltage VR of the ith row, which is applied in the selection period TSE is made equal to that of the voltage supply line Zi in the ith row, the voltages of the electrodes 24A and 24B of the capacitor 24 become equal to each other, so the electric charges written in the capacitor 24 in the preceding frame period TSC are removed.
In addition, although the second transistors 22 and driving transistors 23 of the pixel circuits Di,1 to Di,n are ON, the tone designating current reference voltage VLOW equal to or lower than the reference voltage VSS is applied to the voltage supply line Zi, so the tone designating current IDATA which flows from the voltage supply line Zi to the driving transistors 23 does not flow through the organic EL elements Ei,1 to Ei,n.
FIG. 7 is a circuit diagram showing the states of the electric currents and voltages after the reset period TR in the selection period TSE of the ith row. As shown in FIG. 7, after the reset period TR in the selection period TSE of the ith row, the selection scan driver 5 keeps applying the ON voltage VON to the selection scan line Xi, and the voltage supply driver 6 keeps applying the tone designating current reference voltage VLOW to the voltage supply line Zi. In addition, after the reset period TR in the selection period TSE of the ith row, the current source driver 3 controls the switches s1 to sn to supply the tone designating current IDATA from the current lines Y1 to Yn to the current terminals CT1 to CTn. In the selection period TSE of the ith row, the second transistors 22 of the pixel circuits Di,1 to Di,n in the ith row are ON. Since the second transistors 22 of the pixel circuits Di,1 to Di,n are ON, the voltage is also applied to the gates 23g of the driving transistors 23 of the pixel circuits Di,1 to Di,n, so the driving transistors 23 of the pixel circuits Di,1 to Di,n are turned on. Furthermore, since the first transistors 21 of the pixel circuits Di,1 to Di,n are also ON, the first transistors 21 of the pixel circuits Di,l to Di,n supply the tone designating current IDATA from the voltage supply line Zi to the current lines Y1 to Yn via the drains 23d and sources 23s of the driving transistors 23. In this state, as shown in FIG. 4, the voltage of the current line Yj drops until the tone designating current IDATA becomes steady. Also, although the driving transistors 23 of the pixel circuits Di,1 to Di,n are ON, the low-level tone designating current reference voltage VLOW is applied to the voltage supply line Zi, so no electric current flows from the voltage supply line Zi to the organic EL elements Ei,1 to Ei,n. Therefore, the current value of the tone designating current IDATA flowing through the current lines Y1 to Yn becomes equal to the current value of the electric current IDS between the drain 23d and source 23s of the driving transistor 23. In addition, the level of the voltage between the gate 23g and source 23s of the driving transistor 23 follows the current value of the tone designating current IDATA which flows from the drain 23d to the source 23s. Accordingly, the driving transistor 23 converts the current value of the tone designating current IDATA into the level of the voltage between the gate 23g and source 23s, and electric charges corresponding to the level of the voltage between the gate 23g and source 23s of the driving transistor 23 are held in the capacitor 24. Note that the gate 23g and drain 23d of the driving transistor 23 are connected via the second transistor 22, and the ON resistance of the second transistor 22 upon selection is negligibly low. Therefore, the voltage applied to the gate 23g and the voltage applied to the drain 23d of the driving transistor 23 are substantially equal, so the tone designating current IDATA becomes the electric current IDS which changes on the broken line VTH shown in FIG. 5. That is, when the voltages of the gate 23g and drain 23d of the driving transistor 23 are equal, the voltage VDS between the source 23s and drain 23d is equal to the threshold voltage VTH between the unsaturated and saturated regions.
FIG. 8 is a circuit diagram showing the states of the electric currents and voltages in the non-selection period TNSE of the ith row. As shown in FIG. 8, in the non-selection period TNSE of the ith row, the selection scan driver 5 applies the OFF voltage VOFF to the selection scan line Xi, and the voltage supply driver 6 applies the driving current reference voltage VHIGH to the voltage supply line Zi.
In the non-selection period TNSE of the ith row, the first transistors 21 of the pixel circuits Di,1 to Di,n are OFF. Therefore, the first transistors 21 of the pixel circuits Di,1 to Di,n shut off the tone designating current IDATA flowing through the current lines Y1 to Yn, thereby preventing an electric current from flowing from the voltage supply line Zi to the current lines Y1 to Yn via the driving transistors 23. In addition, since the second transistor 22 of each of the pixel circuits Di,1 to Di,n in the ith row is turned off, the second transistor 22 confines the electric charges in the capacitor 24. In this manner, the second transistor 22 holds the level of the converted voltage between the gate 23g and source 23s of the driving transistor 23, thereby storing the current value of the electric current which flows through the source-to-drain path of the driving transistor 23. In this state, the high-level driving current reference voltage VHIGH by which the source-to-drain voltage VDS of the driving transistor 23 maintains the saturated region is applied to the voltage supply line Zi, and the driving transistor 23 of each of the pixel circuits Di,1 to Di,n is ON. Accordingly, each driving transistor 23 supplies the driving current from the voltage supply line Zi to a corresponding one of the organic EL elements Ei,1 to Ei,n to allow it to emit light at luminance corresponding to the current value of the driving current. In this state, the level of the converted voltage between the gate 23g and source 23s of the driving transistor 23 of each of the pixel circuits Di,1 to Di,n is held by the capacitor 24 so as to be equal to the level of the voltage when the tone designating current IDATA flows through a corresponding one of the current lines Y1 to Yn in the second half of the selection period TSE.
As shown in FIG. 5, a divided voltage VEL of each of the organic EL elements Ei,1 to Ei,n in the non-selection period TNSE is obtained by subtracting, from the driving current reference voltage VHIGH, the voltage VDS on the EL load border line indicated by the alternate long and short dashed line, which is obtained when a driving current (equivalent to IDS shown in FIG. 5) having a current value equal to that of the tone designating current IDATA flows. That is, the voltage difference on the right side of the EL load border line is the divided voltage of one organic EL element. As described above, the divided voltage VEL of the organic EL elements Ei,1 to Ei,n rises as the luminance tone rises. In the non-selection period TNSE, the driving current reference voltage VHIGH is set higher than a voltage obtained by adding the divided voltage VEL when the luminance tone of the organic EL elements Ei,1 to Ei,n is a minimum to the ON voltage VDS between the drain 23d and source 23s of the driving transistor at that time, and higher than a voltage obtained by adding the divided voltage VEL when the luminance tone of the organic EL elements Ei,1 to Ei,n is a maximum to the ON voltage VDS between the drain 23d and source 23s of the driving transistor at that time. Also, in the non-selection period TNSE, the voltage of the source 23s of the driving transistor 23 rises as the voltage VGS between the gate 23g and source 23s, which is held in the selection period TSE rises. Although the capacitor 24 changes the electric charge in the electrode 24B connected to the source 23s accordingly, the voltage VGS between the gate 23g and source 23s is held constant by equally changing the electric charge in the electrode 24A.
As shown in FIG. 5, therefore, between the drain 23d and source 23s of the driving transistor 23 in the non-selection period TNSE is always applied a saturated region voltage, and the current value of the driving current which flows through each of the organic EL elements Ei,1 to Ei,n in the non-selection period TNSE is made equal to the current value of the tone designating current IDATA by the electric charges held between the gate 23g and source 23s in the selection period TSE. Also, as shown in FIG. 4, the voltage of the pixel electrodes 51 of the organic EL elements Ei,1 to Ei,n in the non-selection period TNSE rises as the luminance tone rises. This increases the voltage difference between the pixel electrodes 51 and the common electrode as a cathode, and increases the luminance of the organic EL elements Ei,1 to Ei,n.
As described above, the luminance (the unit is nit.) of the organic EL elements Ei,1 to Ei,n is uniquely determined by the current value of the tone designating current IDATA which flows through the pixel circuits Di,1 to Di,n in the selection period TSE.
A method of driving the organic EL display panel 2 by the current source driver 3, selection scan driver 5, voltage supply driver 6, and switches S1 to Sn, and the display operation of the organic EL display 1 will be described below.
As shown in FIG. 4, the selection scan driver 5 applies the ON voltage VON in order from the selection scan line X1 in the first row to the selection scan line Xm in the mth row (the selection scan line Xm in the mth row is followed by the selection scan line X1 in the first row), thereby selecting these selection scan lines. In synchronism with this selection by the selection scan driver 5, the voltage supply driver 6 applies the tone designating current reference voltage VLOW in order from the voltage supply line Z1 in the first row to the voltage supply line Zm in the mth row (the voltage supply line Zm in the mth row is followed by the voltage supply line Z1 in the first row), thereby selecting these voltage supply lines. In the selection period TSE of each row, the current source driver 3 controls the current terminals CT1 to CTn to generate the tone designating current IDATA having a current value corresponding to the image signal.
Also, at the start of the selection period TSE of each row (at the end of the selection period TSE of the preceding row), the switching signal Φ changes from low level to high level, the switching signal inv.Φ changes from high level to low level, and the reset voltage VR which removes the electric charges stored in the current lines Y1 to Yn and the electric charges stored in the pixel electrodes 51 via the first transistors 21 is applied. In the selection period TSE of each row (at the end of the reset period TR of each row), the switching signal Φ changes from high level to low level, and the switching signal inv.Φ changes from low level to high level. In the reset period TR in the initial part of the selection period TSE, therefore, the switches S1 to Sn allow the tone designating current IDATA to flow between the current terminals CT1 to CTn and current lines Y1 to Yn, and shut down the application of the reset voltage VR to the current lines Y1 to Yn. After the reset period TR in the selection period TSE, the switches S1 to Sn shut off the flow of the electric current between the current terminals CT1 to CTn and current lines Y1 to Yn, and allow the application of the reset voltage VR to the current lines Y1 to Yn.
The current value of the tone designating current IDATA decreases as the luminance tone lowers. In this state, the voltages of the current lines Y1 to Yn and pixel electrodes 51 approximate to the tone designating current reference voltage VLOW, i.e., to the reset voltage VR. Also, if the tone designating current IDATA having a large current value flows in the selection period TSE of the preceding row or of the preceding frame period TSC, the voltage of the pixel electrodes 51 become much lower than the reset voltage VR via the current lines Y1 to Yn and first transistors 21.
If, therefore, no reset voltage is applied to the current lines Y1 to Yn without forming the switches S1 to Sn, and the tone designating current IDATA having a low luminance tone and low current value is to be kept supplied to the ith row, the amount of electric charges to be modulated is large because the electric charges of the current lines Y1 to Yn, which are stored in accordance with the tone designating current IDATA having a large current value in the selection period TSE of the (i−1)th row are held in the parasitic capacitances of the current lines Y1 to Yn. Accordingly, it takes a long time to obtain a desired current value of the tone designating current IDATA.
Likewise, if no reset voltage is applied to the pixel electrodes 51 in the selection period without forming the switches S1 to Sn, and the tone designating current IDATA having a low luminance tone and low current value is to be kept supplied in the next frame period TSC, the amount of electric charges to be modulated are large because the electric charges of the pixel electrodes 51 in the ith row, which are stored in accordance with the tone designating current IDATA having a large current value in the selection period TSE of the frame period TSC before the next frame period TSC are held in the parasitic capacitances of the pixel electrodes 51 in the ith row. Accordingly, it takes a long time to obtain a desired current value of the tone designating current IDATA.
In the selection period TSE, therefore, no sufficient electric charges can be held so that the required voltage is obtained between the gate 23g and source 23s of the driving transistor 23. As a consequence, the driving current in the non-selection period TNSE becomes different from the tone designating current IDATA, and this makes accurate tone display impossible.
Since, however, the switches S1 to Sn which apply the reset voltage VR in the reset period TR are provided, the electric charges stored in the current lines Y1 to Yn and the electric charges stored in the pixel electrodes 51 via the first transistors 21 can be rapidly removed. Accordingly, the voltage between the gate 23g and source 23s of the driving transistor 23 can be rapidly set to a voltage by which the tone designating current IDATA having a low luminance tone and low current value flows. Since this makes high-speed display possible, images particularly excellent in motion image characteristics can be displayed.
FIG. 9 is a timing chart showing, from above, the voltage of the selection scan line X1, the voltage of the voltage supply line Z1, the switching signal inv.Φ, the switching signal Φ, the current value of the current terminal CTj, the current value of an electric current which flows through the driving transistor 23 of the pixel circuit Di,j, the voltage of the pixel electrode 51 of the organic EL element Ei,j, and the current value of an electric current which flows through the organic EL element Ei,j. Referring to FIG. 9, the abscissa represents the common time.
As shown in FIGS. 6 and 9, when the selection scan driver 5 applies the ON voltage VON to the selection scan line Xi in the ith row (i.e., in the selection period TSE of the ith row), the OFF voltage VOFF is applied to the other selection scan lines X1 to Xm (except for Xi). In the selection period TSE of the ith row, therefore, the first and second transistors 21 and 22 of the pixel circuits Di,1 to Di,n in the ith row are ON, and the first and second transistors 21 and 22 of the pixel circuits D1,1 to Dm,n (except for Di,1 to Di,n) in the other rows are OFF.
As described above, in the selection period TSE of the ith row, the tone designating current reference voltage VLOW is applied to the voltage supply line Zi, and the second transistors 22 of the pixel circuits Di,1 to Di,n in the ith row are ON. Accordingly, the voltage is also applied to the gates 23g of the driving transistors 23 of the pixel circuits Di,1 to Di,n in the ith row, so the driving transistors 23 are turned on.
In the reset period TR in the initial part of the selection period TSE of the ith row, the transistors 32 of the switches S1 to Sn are turned on. Therefore, the voltage supply line Zi is electrically connected to the reset input terminal 41 via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n and the current lines Y1 to Yn. In this state, the voltage applied from the voltage supply line Zi to the reset input terminal 41 via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n and the current lines Y1 to Yn is equal to the reset voltage VR (=tone designating current reference voltage VLOW) which is equal to or lower than the reference voltage VSS. Accordingly, the voltage of the pixel electrodes 51 of the organic EL elements Ei,1 to Ei,n is also equal to the reset voltage VR. In addition, since the reset voltage VR is applied to the current lines Y1 to Yn, the electric charges stored in the parasitic capacitances of the current lines Y1 to Yn and the electric charges stored in the parasitic capacitances of the pixel circuits Di,1 to Di,n including the pixel electrodes 51 are removed, so the voltage of these components becomes equal to the reset voltage VR. As a consequence, the organic EL elements Ei,1 to Ei,n stop emitting light immediately after the start of the reset period TR of the ith row.
As shown in FIGS. 7 and 9, in the second half of the selection period TSE after the reset period TR, the ON voltage VON is applied to the selection scan line Xi in the ith row, and the tone designating current reference voltage VLOW is applied to the voltage supply line Zi in the ith row. Therefore, the first transistors 21, second transistors 22, and driving transistors 23 of the pixel circuits Di,1 to Di,n in the ith row are ON. After the reset period TR in the selection period TSE, the transistors 31 of the switches S1 to Sn are turned on, so the switches S1 to Sn allow an electric current to flow between the current terminals CT1 to CTn and current lines Y1 to Yn. As a consequence, the current terminals CT1 to CTn are electrically connected to the voltage supply line Zi in the ith row. In this state, the current source driver 3 supplies the tone designating current IDATA from the voltage supply line Zi to the current terminals CT1 to CTn via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n the current lines Y1 to Yn, and the switches S1 to Sn. Until the end of the selection period TSE of the ith row, the current source driver 3 controls the current value of the tone designating current IDATA supplied to the current lines Y1 to Yn such that the current value is held constant in accordance with the image signal.
In the second half of the selection period TSE of the ith row, the tone designating current IDATA flows along the voltage supply line Zi→the path between the drain 23d and source 23s of the driving transistor 23 of each of the pixel circuits Di,1 to Di,n→the path between the drain 21d and source 21s of the first transistor 21 of each of the pixel circuits Di,1 to Di,n→the current lines Y1 to Yn→the transistors 31 of the switches S1 to Sn→the current terminals CT1 to CTn of the current source driver 3. In the selection period TSE of the ith row, therefore, the voltage applied from the voltage supply line Zi to the current terminals CT1 to CTn via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n and the current lines Y1 to Yn becomes steady.
That is, since the voltage applied from the voltage supply line Zi in the ith row to the current terminals CT1 to CTn becomes steady, the voltage having a level corresponding to the current value of the tone designating current IDATA which flows through the driving transistor 23 is applied between the gate 23g and source 23s of the driving transistor 23, so electric charges corresponding to the level of this voltage between the gate 23g and source 23s of the driving transistor 23 is held in the capacitor 24. Consequently, the current value of the tone designating current IDATA which flows through the driving transistor 23 of each of the pixel circuits Di,1 to Di,n in the ith row is converted into the level of the voltage between the gate 23g and source 23s of the driving transistor 23.
In the reset period TR of the ith row as described above, the reset voltage VR is applied to the current lines Y1 to Yn. Therefore, the voltage applied from the voltage supply line Zi to the reset input terminal 41 via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n and the current lines Y1 to Yn can be made steady. Accordingly, even if a weak tone designating current IDATA flows through the current lines Y1 to Yn after the reset period TR of the ith row, electric charges corresponding to the tone designating current IDATA can be rapidly held in the capacitors 24 of the pixel circuits Di,1 to Di,n.
As described above, the current value of the electric current which flows between the drain 23d and source 23s of the driving transistor 23 of each of the pixel circuits Di,1 to Di,n in the ith row and the level of the voltage between the source 23s and gate 23g are overwritten from those of the preceding frame period TSC. In the selection period TSE Of the ith row, therefore, the magnitude of the electric charges which are held in the capacitor 24 of each of the pixel circuits Di,1 to Di,n in the ith row is overwritten from that of the preceding frame period TSC.
The potential at arbitrary points in the paths from the driving transistors 23 of the pixel circuits Di,1 to Di,n to the current lines Y1 to Yn via the first transistors 21 changes in accordance with, e.g., the internal resistances of the transistors 21, 22, and 23, which change with time. In this embodiment, however, in the selection period TSE, the current source driver 3 forcedly supplies the tone designating current IDATA from the driving transistors 23 of the pixel circuits Di,1 to Di,n to the current lines Y1 to Yn via the first transistors 21. Therefore, even if the internal resistances of the transistors 21, 22, and 23 change with time, the tone designating current IDATA takes a desired current value.
Also, in the selection period TSE of the ith row, the common electrode of the organic EL elements Ei,1 to Ei,n in the ith row is at the reference voltage VSS, and the voltage supply line Zi is at the tone designating current reference voltage VLOW which is equal to or lower than the reference voltage VSS. As a consequence, a reverse bias voltage is applied to the organic EL elements Ei,1 to Ei,n in the ith row. Accordingly, no electric current flows through the organic EL elements Ei,1 to Ei,n in the ith row, so the organic EL elements Ei,1 to Ei,n do not emit light.
Subsequently, as shown in FIGS. 8 and 9, at the end time of the selection period TSE of the ith row (at the start time of the non-selection period TNSE of the ith row), a signal output from the selection scan driver 5 to the selection scan line Xi changes from the high-level ON voltage VON to the low-level OFF voltage VOFF. That is, the selection scan driver 5 applies the OFF voltage VOFF to the gate 21g of the first transistor 21 and the gate 22g of the second transistor 22 of each of the pixel circuits Di,1 to Di,n in the ith row.
In the non-selection period TNSE Of the ith row, therefore, the first transistors 21 of the pixel circuits Di,1 to Di,n in the ith row are turned off to prevent the electric current from flowing from the voltage supply line Zi to the current lines Y1 to Yn. In addition, in the non-selection period TNSE of the ith row, when the second transistors 22 of the pixel circuits Di,1 to Di,n in the ith row are turned off, the electric charges held in the capacitors 24 in the immediately preceding selection period TSE of the ith row are confined by the second transistors 22. Accordingly, the driving transistor 23 of each of the pixel circuits Di,1 to Di,n in the ith row is kept ON in the non-selection period TNSE. That is, in each of the pixel circuits Di,1 to Di,n in the ith row, the voltage VGS between the gate 23g and source 23s of the driving transistor 23 in the non-selection period TNSE becomes equal to the voltage VGS between the gate 23g and source 23s of the driving transistor 23 in the immediately preceding selection period TSE, i.e., the capacitor 24 in which the electric charges on the side of the electrode 24A are held by the second transistor 22 holds the voltage VGS between the gate 23g and source 23s of the driving transistor 23.
Also, in the non-selection period TNSE of the ith row, the voltage supply driver 6 applies the driving current reference voltage VHIGH to the voltage supply line Zi in the ith row. In the non-selection period TNSE, the common electrode of the organic EL elements Ei,1 to Ei,n in the ith row is at the reference voltage VSS, and the voltage supply line Zi in the ith row is at the driving current reference voltage VHIGH which is higher than the reference voltage VSS, so the driving transistors 23 of the pixel circuits Di,1 to Di,n in the ith row are ON. As a consequence, a forward bias voltage is applied to the organic EL elements Ei,1 to Ei,n. In the pixel circuits Di,1 to Di,n, therefore, a driving current flows from the voltage supply line Zi to the organic EL elements Ei,1 to Ei,n via the driving transistors 23, and thus the organic EL elements Ei,1 to Ei,n emit light.
More specifically, in the pixel circuit Di,j in the non-selection period TNSE of the ith row, the first transistor 21 electrically shuts off the path between the current line Yj and driving transistor 23, and the second transistor 22 confines the electric charges in the capacitor 24. In this manner, the level of the voltage, which is converted in the selection period TSE, between the gate 23g and source 23s of the driving transistor 23 is held, and a driving current having a current value corresponding to the level of this voltage held between the gate 23g and source 23s is supplied to the organic EL element Ei,j by the driving transistor 23.
In this state, the current value of the driving current which flows through the organic EL elements Ei,1 to Ei,n in the selection period TSE of the ith row is equal to the current value of the electric current which flows through the driving transistors 23 of the pixel circuits Di,1 to Di,n, and therefore equal to the current value of the tone designating current IDATA which flows through the driving transistors 23 of the pixel circuits Di,1 to Di,n in the selection period TSE. As described above, in the selection period TSE, the current value of the tone designating current IDATA which flows through the driving transistors 23 of the pixel circuits Di,1 to Di,n is a desired current value. Therefore, a driving current having a desired current value can be supplied to the organic EL elements Ei,1 to Ei,n, so the organic EL elements Ei,1 to Ei,n can emit light at a desired tone luminance.
In the reset period TR of the (i+1)th row after the selection period TSE of the ith row, as in the reset period TR of the ith row, the transistors 31 of the switches S1 to Sn are turned off, and the transistors 32 of the switches S1 to Sn are turned on. Accordingly, in the reset period TR of the (i+1)th row, the tone designating current IDATA does not flow through any of the current lines Y1 to Yn, but the reset voltage VR is applied to all the current lines Y1 to Yn, the pixel electrodes 51 in the (i+1)th row, the electrodes 24B of the capacitors 24 in the (I+1)th row, and the sources 23s of the driving transistors 23 in the (i+1)th row. After the reset period TR in the selection period TSE of the (i+1)th row, as in the case of the ith row, the selection scan driver 5 selects the selection scan line Xi+1 in the (i+1)th row, so the tone designating current IDATA flows from the voltage supply line Zi to the current terminals CT1 to CTn via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n, the current lines Y1 to Yn, and the switches Di,1 to Di,n.
As described above, in the reset period TR, the reset voltage VR is forcedly applied to, e.g., the current lines Y1 to Yn and the pixel electrodes 51. Therefore, the charge amount of the parasitic capacitances of the current lines Y1 to Yn and the like approximates to the charge amount in a steady state in which a small electric current flows. Accordingly, even when the electric current which flows through the current lines Y1 to Yn after the reset period TR of the (i+1)th row is weak, a steady state can be immediately obtained.
In this embodiment as described above, the current value of the driving current which flows through the organic EL elements E1,1 to Em,n in the non-selection period TNSE is represented by the current value of the tone designating current IDATA after the reset period TR of the selection period TSE. Therefore, even when variations are produced in characteristics of the driving transistors 23 of the pixel circuits D1,1 to Dm,n, no variations are produced in luminance of the organic EL elements E1,1 to Em,n if the current value of the tone designating current IDATA remains the same for all the pixel circuits D1,1 to Dm,n. That is, this embodiment can suppress planar variations by which pixels have different luminance values even though luminance tone signals having the same level are output to these pixels. Accordingly, the organic EL display 1 of this embodiment can display high-quality images.
The tone designating current IDATA is very weak because it is equal to the current value of the electric current which flows through the organic EL elements E1,1 to Em,n in accordance with the luminance of the organic EL elements E1,1 to Em,n which emit light. The wiring capacitances of the current lines Y1 to Yn delay the tone designating current IDATA which flows through the current lines Y1 to Yn. If the selection period TSE is short, therefore, electric charges corresponding to the tone designating current IDATA cannot be held in the gate-to-source path of the driving transistor 23. In this embodiment, however, the reset voltage VR is forcedly applied to the current lines Y1 to Yn in the reset period TR of each row. Therefore, even if the tone designating current IDATA is weak or the selection period TSE is short, electric charges corresponding to the tone designating current IDATA can be held in the gate-to-source path of the driving transistor 23 within the selection period TSE.
Also, in this embodiment, the data driving circuit 7 applies the reset voltage VR to the current lines Y1 to Yn in the selection period TSE. Therefore, the first transistor 21 has both the function of a switching element which loads the reset voltage VR into each of the pixel circuits D1,1 to Dm,n, and the function of a switching element which loads the tone designating current IDATA into each of the pixel circuits D1,1 to Dm,n. This makes it unnecessary to form any switching TFT, which loads a blanking signal into a pixel circuit as in the conventional device (Jpn. Pat. Appln. KOKAI Publication No. 2000-221942), in the pixel circuits D1,1 to Dm,n in addition to the first transistors 21. Accordingly, the number of transistors necessary for the pixel circuits D1,1 to Dm,n does not increase. When the organic EL elements E1,1 to Em,n are formed on the same surface as the pixel circuits D1,1 to Dm,n, therefore, the aperture ratio of the pixels P1,1 to Pm,n does not decrease.
Second Embodiment
FIG. 10 is a block diagram showing an organic EL display 101 according to the second embodiment to which the organic EL display of the present invention is applied. In FIG. 10, the same reference numerals and symbols as in the organic EL display 1 of the first embodiment denote the same parts in the organic EL display 101, and an explanation thereof will be omitted.
Similar to the organic EL display 1 shown in FIG. 1, the organic EL display 101 includes an organic EL display panel 2, scan driving circuit 9, and data driving circuit 107. The organic EL display panel 2 and scan driving circuit 9 are the same as the organic EL display panel 2 and scan driving circuit 9 of the first embodiment. The data driving circuit 107 is different from the data driving circuit 7 of the first embodiment.
The data driving circuit 107 includes n current terminals DT1 to DTn, a current control driver 103 which supplies a pull current IL1 to the current terminals DT1 to DTn, first current mirror circuits M11 to Mn1 and second current mirror circuits M12 to Mn2 which convert the pull current IL1 flowing through the current terminals DT1 to DTn into a tone designating current IDATA, and switches T1 to Tn interposed between current lines Y1 to Yn, the first current mirror circuits M11 to Mn1, and the second current mirror circuits M12 to Mn2.
An 8-bit digital tone image signal is input to the current control driver 103. This digital tone image signal loaded into the current control driver 103 is converted into an analog signal by an internal D/A converter of the current control driver 103. The driver 103 generates the pull current IL1 having a current value corresponding to the analog image signal at the current terminals DT1 to DTn. The driver 103 supplies the pull current IL1 from the first current mirror circuits M11 to Mn1 formed for individual rows to the current terminals DT1 to DTn. In accordance with the pull current IL1, the current control driver 103 supplies the tone designating current IDATA from driving transistors 23 in the individual rows to the second current mirror circuits M12 to Mn2 via the current lines Y1 to Yn.
The operation timings of the current control driver 103 are the same as those of the current source driver 3 of the first embodiment. That is, the current control driver 103 controls the current value of the pull current IL1 at the current terminals DT1 to DTn in each selection period TSE of each row in accordance with the image signal, and makes the current value of the pull current IL1 steady in a period from the end of each reset period TR to the end of the corresponding selection period TSE. The pull current IL1 supplied by the current control driver 103 is larger than and proportional to the tone designating current IDATA supplied by the current source driver 3 of the first embodiment.
The first current mirror circuits M11 to Mn1 and second current mirror circuits M12 to Mn2 convert the pull current IL1 which flows through the current terminals DT1 to DTn into the tone designating current IDATA at a predetermined conversion ratio. Each of the first current mirror circuits M11 to Mn1 is made up of two P-channel MOS transistors 61 and 62. The transistors 61 and 62 can be fabricated by the same steps as the transistors 21 to 23 of each of pixel circuits D1,1 to Dm,n. Each of the second current mirror circuits M12 to Mn2 is made up of two N-channel MOS transistors 63 and 64. The transistors 63 and 64 can be partially fabricated by the same steps as the transistors 21 to 23 of each of the pixel circuits D1,1 to Dm,n.
In the first current mirror circuits M11 to Mn1, the gates and drains of the transistors 61 and the gates of the transistors 62 are connected to the current terminals DT1 to DTn. The sources of the transistors 61 and 62 are connected to a reset input terminal 41 which outputs a reset voltage VR as a ground voltage.
In the second current mirror circuits M12 to Mn2, the gates and drains of the transistors 63 and the gates of the transistors 64 are connected together to the drains of the transistors 62. The sources of the transistors 63 and 64 are connected to a constant-voltage input terminal 45 to which a negative voltage VCC is applied, and the drains of the transistors 64 are connected to the sources of transistors 34 of the switches T1 to Tn (to be described later). In each of the first current mirror circuits M11 to Mn1, the channel resistance of the transistor 61 is lower than that of the transistor 62. In each of the second current mirror circuits M12 to Mn2, the channel resistance of the transistor 63 is lower than that of the transistor 64.
Each of the switches T1 to Tn has an N-channel MOS transistor 33 and the N-channel MOS transistor 34. The transistors 33 and 34 can be fabricated by the same steps as the transistors 21 to 23 of each of the pixel circuits D1,1 to Dm,n. An example of the switch Tj will be explained below. The gate of the transistor 34 of the switch Tj is connected to a switching signal input terminal 43, and thus a switching signal inv.Φ is input to the gate of the transistor 34. Also, the gate of the transistor 33 is connected to a switching signal input terminal 42, and thus a switching signal Φ is input to the gate of the transistor 33. The drains of the transistors 33 and 34 are connected to the current line Yj, the source of the transistor 33 is connected to the source of the transistor 61 of the first current mirror circuit Mi1 and the reset input terminal 41, and the source of the transistor 34 is connected to the drain of the transistor 64 of the second current mirror circuit Mi2.
In this arrangement, when the switching signal Φ is at high level and the switching signal inv.Φ is at low level, the transistor 33 is turned on, and the transistor 34 is turned off. The switching signals Φ and inv.Φ have the same waveforms as in FIG. 4 of the first embodiment. Accordingly, the switches T1 to Tn switch the state in which the tone designating current IDATA obtained by modulating the current value of the pull current IL1 by the first current mirror circuits M11 to Mn1 and second current mirror circuits M12 to Mn2 is supplied to the driving transistors 23 and current lines Y1 to Yn, and the state in which the reset voltage VR is applied to the current lines Y1 to Yn.
When the current control driver 103 supplies the pull current IL1 to the current terminal DTj, an electric current which flows through the drain-to-source path of the transistor 62 in the first current mirror circuit Mj1 has a value obtained by multiplying the ratio of the channel resistance of the transistor 62 to that of the transistor 61 by the current value of the pull current IL1 in the drain-to-source path of the transistor 61. In the second current mirror circuit Mj2, an electric current which flows through the drain-to-source path of the transistor 64 has a value obtained by multiplying the ratio of the channel resistance of the transistor 64 to that of the transistor 63 by the current value of an electric current in the drain-to-source path of the transistor 63. The current value of the electric current in the drain-to-source path of the transistor 63 matches the electric current which flows through the drain-to-source path of the transistor 62. Therefore, the current value of the tone designating current IDATA is obtained by multiplying the ratio of the channel resistance of the transistor 64 to that of the transistor 63 by the value which is obtained by multiplying the ratio of the channel resistance of the transistor 62 to that of the transistor 61 by the current value of the pull current IL1 in the drain-to-source path of the transistor 61.
As described above, the first current mirror circuits M11 to Mn1 and second current mirror circuits M12 to Mn2 convert the pull current IL1 which flows through the current terminals DT1 to DTn into the tone designating current IDATA. Since the tone designating current IDATA flows through the output sides of the second current mirror circuits M12 to Mn2, i.e., the drains of the transistors 64, these drains of the transistors 64 of the second current mirror circuits M12 to Mn2 are equivalent to the current terminal CTj of the current source driver 3 of the first embodiment. That is, an arrangement obtained by combining the first current mirror circuits M11 to Mn1, second current mirror circuits M12 to Mn2, and current control driver 103 is equivalent to the current source driver 3 of the first embodiment.
In the first embodiment, the reset voltage VR is at the same level as the tone designating current reference voltage VLOW. In the second embodiment, however, the reset voltage VR is set at 0 [V]. Therefore, when a voltage VSS is set at the ground voltage, no voltage difference is produced between pixel electrodes 51 as the anodes of the organic EL elements E1,1 to Em,n and the common electrode as the cathode. As a consequence, electric charges stored in the pixel electrodes 51 can be easily removed.
In order for the switches T1 to Tn to perform the switching operation, as in the first embodiment, the switching signal Φ is input to the switching signal input terminal 42, and the switching signal inv.Φ is input to the switching signal input terminal 43. The relationship between the timings of the switching signals Φ and inv.Φ and the selection timings of a selection scan driver 5 and voltage supply driver 6 is the same as in the first embodiment. Also, the operation timings of the selection scan driver 5 and voltage supply driver 6 in the second embodiment are the same as in the first embodiment.
In the second embodiment, as in the first embodiment, in the reset period TR of the former period in the selection period TSE of the ith row, the transistors 33 of the switches T1 to Tn are turned on, so a voltage supply line Zi is electrically connected to the reset input terminal 41 via the driving transistors 23 and first transistors 21 of the pixel circuits Di,1 to Di,n and the current lines Y1 to Yn.
Also, in the reset period TR of the ith row, the reset voltage VR is applied to the current lines Y1 to Yn and pixel electrodes 51, so the electric charges stored in the parasitic capacitances of the current lines Y1 to Yn and the electric charges stored in the parasitic capacitances of the pixel electrodes 51 can be rapidly removed. Accordingly, even when the weak tone designating current IDATA flows through the current lines Y1 to Yn after the reset period TR of the ith row, electric charges corresponding to the tone designating current IDATA can be rapidly held in capacitors 24 of the pixel circuits Di,1 to Di,n.
In addition, in a non-selection period TNSE, the current value of a driving current which flows through the organic EL elements E1,1 to Em,n is represented by the current value of the tone designating current IDATA after the reset period TR of each selection period TSE. Therefore, even if variations are produced in Characteristics of the driving transistors 23 of the pixel circuits D1,1 to Dm,n, no variations are produced in driving current because the tone designating current IDATA is forcedly supplied to the driving transistors 23. As a consequence, no variations are produced in luminance of the organic EL elements E1,1 to Em,n.
Furthermore, since the first current mirror circuits M11 to Mn1 and second current mirror circuits M12 to Mn2 are formed, the current value of the tone designating current IDATA of the current lines Y1 to Yn is proportional to and smaller than the pull current IL1 at the current terminals DT1 to DTn. Accordingly, even if the pull current IL1 at the current terminals DT1 to DTn is unexpectedly reduced by a leakage current produced in the current control driver 103 or the like, the tone designating current IDATA of the current lines Y1 to Yn does not largely reduce. That is, even a decrease in output from the current control drive 103 caused by a current leak has no large influence on the tone designating current IDATA Of the current lines Y1 to Yn, so the luminance of the organic EL elements E1,1 to Em,n does not largely decrease.
In the second embodiment, the data driving circuit 107 can well generate the tone designating current IDATA even when the current control driver 103 cannot generate a weak electric current close to the tone designating current IDATA matching the light emission characteristics of the organic EL elements.
The data driving circuit 107 applies the reset voltage VR to the current lines Y1 to Yn in the selection period TSE in the second embodiment as well. Therefore, the first transistor 21 has both the function of a switching element which loads the reset voltage VR into each of the pixel circuits D1,1 to Dm,n, and the function of a switching element which loads the tone designating current IDATA into each of the pixel circuits D1,1 to Dm,n. Accordingly, the number of transistors necessary for the pixel circuits D1,1 to Dm,n does not increase. When the organic EL elements E1,1 to Em,n are formed on the same surface as the pixel circuits D1,1 to Dm,n, therefore, the aperture ratio of the pixels P1,1 to Pm,n does not decrease.
Third Embodiment
FIG. 11 is a block diagram showing an organic EL display 201 according to the third embodiment to which the organic EL display of the present invention is applied. In FIG. 11, the same reference numerals and symbols as in the organic EL display 1 of the first embodiment denote the same parts in the organic EL display 201, and an explanation thereof will be omitted.
Similar to the organic EL display 1, the organic EL display 201 includes an organic EL display panel 2, scan driving circuit 9, and data driving circuit 207. The organic EL display panel 2 and scan driving circuit 9 are the same as the organic EL display panel 2 and scan driving circuit 9 of the first embodiment. The data driving circuit 207 is different from the data driving circuit 7 of the first embodiment.
The data driving circuit 207 includes a current control driver 203 which has n current terminals FT1 to FTn and supplies a push current IL2 to the current terminals FT1 to FTn, current mirror circuits M1 to Mn for converting the push current IL2 flowing through the current terminals FT1 to FTn, and switches S1 to Sn interposed between current lines Y1 to Yn and the current mirror circuits M1 to Mn.
In the second embodiment, the current control driver 103 supplies the pull current IL1 from the current mirror circuits M1 to Mn to the current terminals DT1 to DTn. In the third embodiment, the current control driver 203 supplies the push current IL2 from the current terminals FT1 to FTn to the current mirror circuits M1 to Mn.
Each of the current mirror circuits M1 to Mn is made up of two N-channel MOS transistors 161 and 162. The transistors 161 and 162 can be fabricated by the same steps as transistors 21 to 23 of pixel circuits D1,1 to Dm,n.
In each of the current mirror circuits M1 to Mn, the gate and drain of the transistor 161 and the gate of the transistor 162 are connected together, and the sources of the transistors 161 and 162 are connected to a constant-voltage input terminal 45. A constant voltage VCC is applied to the constant-voltage input terminal 45. The level of the constant voltage VCC is lower than a tone designating current reference voltage VLOW and reference voltage VSS. When the reference voltage VSS or tone designating current reference voltage VLOW is 0 [V] as in the first embodiment, the constant voltage VCC is a negative voltage.
An example of the switch Sj will be explained below. The switch Sj is made up of N-channel field-effect transistors 31 and 32. The gate of the transistor 31 is connected to a switching signal input terminal 43, and thus a switching signal inv.Φ is input to the gate of the transistor 31. Also, the gate of the transistor 32 is connected to a switching signal input terminal 42, and thus a switching signal Φ is input to the gate of the transistor 32. The drain of the transistor 31 is connected to the current line Yj, and the source of the transistor 31 is connected to the drain of the transistor 162. The drain of the transistor 32 is connected to the current line Yj. The source of the transistor 32 is connected to a reset input terminal 41, and thus a reset voltage VR as a constant voltage is applied to the source of the transistor 32. In this arrangement, when the switching signal Φ is at high level and the switching signal inv.Φ is at low level, the transistor 32 is turned on, and the transistor 31 is turned off. When the switching signal Φ is at low level and the switching signal inv.Φ is at high level, the transistor 31 is turned on, and the transistor 32 is turned off. The transistors 31 and 32 can be fabricated by the same steps as the transistors 21 to 23 of the pixel circuits D1,1 to Dm,n. The reset voltage VR is preferably 0 [V] in order to completely discharge, e.g., the electric charges stored in the parasitic capacitances of the current lines Y1 to Yn and the electric charges stored in the parasitic capacitances of pixel electrodes 51.
The current control driver 203 controls the current value of the push current IL2 at the current terminals FT1 to FTn in accordance with the image signal in each selection period TSE of each row, and holds the magnitude of the push current IL2 constant in a period from the end of each reset period TR to the end of the corresponding selection period TSE. The push current IL2 supplied by the current control driver 203 is larger than and proportional to the tone designating current IDATA supplied by the current source driver 3 of the first embodiment.
The channel resistance of the transistor 161 is lower than that of the transistor 162. Therefore, the current mirror circuits M1 to Mn convert the push current IL2 which flows through the current terminals FT1 to FTn into a tone designating current IDATA. The current value of the tone designating current IDATA is substantially a value obtained by multiplying the ratio of the cannel resistance of the transistor 161 to that of the transistor 162 by the current value of the push current IL2 in the drain-to-source path of the transistor 161. Since the tone designating current IDATA flows through the output sides of the current mirror circuits M1 to Mn, i.e., the drains of the transistors 162, these drains of the transistors 162 of the current mirror circuits M1 to Mn are equivalent to the current terminals CT1 to CTn of the current source driver 3 of the first embodiment. That is, an arrangement obtained by combining the current mirror circuits M1 to Mn and current control driver 203 is equivalent to the current source driver 3 of the first embodiment.
The relationship between the timings of the switching signals Φ and inv.Φ and the selection timings of the selection scan driver 5 and voltage supply driver 6 in this embodiment is the same as in the first embodiment. Also, the operation timings of the selection scan driver 5 and voltage supply driver 6 in the third embodiment are the same as in the first embodiment. Therefore, in the reset period TR of the ith row, the first transistors 21 of the pixel circuits D1,1 to Dm,n are ON in the third embodiment as well. Accordingly, the voltages of the pixel electrodes 51 of organic EL elements Ei,1 to Ei,n, drains 21d of the first transistors 21 in the ith row, electrodes 24B of capacitors 24 in the ith row, sources 23s of the driving transistors 23 in the ith row, and the current lines Y1 to Yn are set in a steady state, thereby removing the electric charges stored in these parasitic capacitances in the preceding selection period TSE. Consequently, the tone designating current IDATA can be rapidly and accurately written in the next selection period TSE.
The data driving circuit 207 applies the reset voltage VR to the current lines Y1 to Yn in the selection period TSE in the third embodiment as well. Therefore, the first transistor 21 has both the function of a switching element which loads the reset voltage VR into each of the pixel circuits D1,1 to Dm,n, and the function of a switching element which loads the tone designating current IDATA into each of the pixel circuits D1,1 to Dm,n. Accordingly, the number of transistors necessary for the pixel circuits D1,1 to Dm,n does not increase. When the organic EL elements E1,1 to Em,n are formed on the same surface as the pixel circuits D1,1 to Dm,n, therefore, the aperture ratio of the pixels P1,1 to Pm,n does not decrease.
Fourth Embodiment
FIG. 12 is a block diagram showing an organic EL display 301 according to the fourth embodiment to which the organic EL display of the present invention is applied. In FIG. 12, the same reference numerals and symbols as in the organic EL display 1 of the first embodiment denote the same parts in the organic EL display 301, and an explanation thereof will be omitted.
Similar to the organic EL display 1, the organic EL display 301 includes an organic EL display panel 2, scan driving circuit 9, and data driving circuit 307. The organic EL display panel 2 and scan driving circuit 9 are the same as the organic EL display panel 2 and scan driving circuit 9 of the third embodiment. The data driving circuit 307 is different from the data driving circuit 7 of the first embodiment.
The data driving circuit 307 includes a current control driver 303, current mirror circuits M1 to Mn, switching elements K1 to Kn, and switching elements W1 to Wn as switches.
The current control driver 303 has n current terminals GT1 to GTn. An 8-bit digital tone image signal is input to the current control driver 303. This digital tone image signal loaded into the current control driver 303 is converted into an analog signal by an internal D/A converter of the current control driver 303. The current control driver 303 generates a push current IL3 having a current value corresponding to the analog image signal at the current terminals GT1 to GTn. The current control driver 303 controls the current value of the push current IL3 at the current terminals GT1 to GTn in each selection period TSE of each row in accordance with the image signal, and holds the current value of the push current IL3 constant in a period from the end of each reset period TR to the end of the corresponding selection period TSE. The push current IL3 supplied by the current control driver 303 is larger than the tone designating current IDATA supplied by the current source driver 3 of the first embodiment, and proportional to a tone designating current IDATA which flows through a transistor 362 (to be described later).
The current mirror circuits M1 to M1 convert the push current IL3 which flows through the current terminals GT1 to GTn into the tone designating current IDATA. Each of the current mirror circuits M1 to Mn has two transistors 361 and 362. In the current mirror circuit Mj, the gate of the transistor 361 is connected to the gate of the transistor 362, and the drain of the transistor 361 is connected to the current terminal and to the gates of the transistors 361 and 362. The drain of the transistor 362 is connected to a current line Yj. The sources of the transistors 361 and 362 are connected to a common voltage terminal 344. A constant voltage VCC is applied to the voltage terminal 344. The level of the constant voltage VCC is lower than a tone designating current reference voltage VLOW and reference voltage VSS. When the reference voltage VSS or tone designating current reference voltage VLOW is 0 [V] as in the first embodiment, the constant voltage VCC is a negative voltage.
The current value of the tone designating current IDATA is substantially a value obtained by multiplying the ratio of the cannel resistance of the transistor 362 to that of the transistor 361 by the current value of the push current IL3 in the drain-to-source path of the transistor 361. That is, an arrangement obtained by combining the current mirror circuits M1 to Mn and current control driver 303 is equivalent to the current source driver.
The drains of the transistors or switching elements W1 to Wn are connected to the current terminals GT1 to GTn and to the drains and gates of the transistors 361 of the current mirror circuits M1 to Mn. The sources of the switching elements W1 to Wn are connected to the voltage terminal 344. The gates of the switching elements W1 to Wn are connected to a switching signal input terminal 42. The switching elements W1 to Wn switch the application of the constant voltage VCC to the drains of the transistors 361 of the current mirror circuits M1 to Mn. Note that the switching elements W1 to Wn may also be incorporated into the current control driver 303.
The relationship between the timings of switching signals and the selection timings of a selection scan driver 5 and voltage supply driver 6 in this embodiment is the same as in the first embodiment.
In the reset period TR in the initial part of the selection period TSE of the ith row, therefore, the transistors W1 to Wn are turned on, so the voltages of the sources and drains of the transistors 361 become equal to each other. Accordingly, after the reset period TR of the selection period TSE, the influence of the parasitic capacitances of the current mirror circuits M1 to Mn on the current lines Y1 to Yn can be removed.
In each of switching elements K1 to Kn, one of the drain and source is connected to a reset input terminal 41, the other of the drain and source is connected to a corresponding one of the current lines Y1 to Yn, and the gate is connected to the switching signal input terminal 42. The switching elements K1 to Kn switch the application of the reset voltage VR to the current lines Y1 to Yn. The reset voltage VR is set at 0 [V]. Note that on the opposite side of the connecting portion between each of the current lines Y1 to Yn and the transistor 362, the other of the drain and source of a corresponding one of the switching elements K1 to Kn may also be connected to a corresponding one of the current lines Y1 to Yn, and the switching elements K1 to Kn may also be formed on the organic EL display panel 2.
In the reset period TR in the initial part of the selection period TSE of the ith row, the switching elements K1 to Kn are turned on, so pixel electrodes 51 and the current lines Y1 to Yn electrically conduct to the reset input terminal 41 to apply the grounded reset voltage VR. Therefore, immediately after the start of the reset period TR of the ith row, it is possible to remove the electric charges stored in the parasitic capacitances of the current lines Y1 to Yn, the electric charges stored in the parasitic capacitances of the pixel electrodes 51, the electric charges stored in the parasitic capacitances of electrodes 24B of capacitors 24, and the electric charges stored in the parasitic capacitances of the sources of driving transistors 23. Accordingly, the tone designating current IDATA having a very small current value can be accurately and rapidly supplied. After the reset period TR, the switching elements K1 to Kn and W1 to Wn are turned off, and an electric current having a current value corresponding to the tone flows through the current terminals GT1 to GTn of the current control driver 303. Consequently, the tone designating current IDATA modulated by the current mirror circuits M1 to Mn flow through the current lines Y1 to Yn and driving transistor 23.
The data driving circuit 307 applies the reset voltage VR to the current lines Y1 to Yn in the selection period TSE in the fourth embodiment as well. Therefore, a first transistor 21 has both the function of a switching element which loads the reset voltage VR into each of the pixel circuits D1,1 to Dm,n, and the function of a switching element which loads the tone designating current IDATA into each of the pixel circuits D1,1 to Dm,n. Accordingly, the number of transistors necessary for the pixel circuits D1,1 to Dm,n does not increase. When organic EL elements E1,1 to Em,n are formed on the same surface as the pixel circuits D1,1 to Dm,n, therefore, the aperture ratio of the pixels P1,1 to Pm,n does not decrease.
The present invention is not limited to the above embodiments, and various improvements and design changes can be made without departing from the spirit and scope of the present invention.
For example, an organic EL element is used as a light-emitting element in each of the above embodiments. However, another light-emitting element having rectification characteristics may also be used. That is, it is also possible to use a light-emitting element in which no electric current flows if a reverse bias voltage is applied and an electric current flows if a forward bias voltage is applied, and which emits light at luminance corresponding to the current value of the flowing electric current. An example of the light-emitting element having rectification characteristics is an LED (Light-Emitting Diode).
In addition, the tone designating current reference voltage VLOW of the voltage supply driver 6 may also be positioned on the right side of the EL load border line corresponding to the maximum luminance tone shown in FIG. 4, provided that a portion or the whole of the tone designating current IDATA does not flow through the organic EL elements in the selection period TSE.

## Claims

1. A display device comprising:
a plurality of selection scan lines;
a plurality of current lines;
a selection scan driver which sequentially selects said plurality of selection scan lines in each selection period;
a data driving circuit which applies a reset voltage to said plurality of current lines in a first part of the selection period, and supplies a designating current having a current value corresponding to an image signal to said plurality of current lines in a second part of the selection period after applying the reset voltage in the selection period; and
a plurality of pixel circuits which are connected to said plurality of selection scan lines and said plurality of current lines, and supply a driving current having a current value corresponding to the current value of the designating current which flows through said plurality of current lines;
wherein in the selection period, each of said plurality of pixel circuits loads the designating current which flows through said plurality of current lines, and stores a level of a voltage converted in accordance with the current value of the designating current, and after the selection period, each of said plurality of pixel circuits shuts off the designating current which flows through said plurality of current lines, and supplies a driving current corresponding to the level of the voltage converted in accordance with the designating current.

2. An apparatus according to claim 1, wherein said data driving circuit comprises:
a switch which switches to a state in which the reset voltage is applied to said plurality of current lines in the first part of the selection period; and
a current source driver which supplies the designating current having the current value corresponding to the image signal after the reset voltage is applied by the switch within the selection period.

3. An apparatus according to claim 1, further comprising a plurality of light-emitting elements which are arranged at intersections of said plurality of selection scan lines and said plurality of current lines, emit light at luminance corresponding to a current value of a driving current, and each have two electrodes one of which is connected to a corresponding one of said plurality of pixel circuits.

4. An apparatus according to claim 3, wherein the reset voltage applied by the data driving circuit is set equal to or lower than a voltage of the other electrode of the light-emitting element.

5. An apparatus according to claim 1, further comprising:
a plurality of voltage supply lines; and
a voltage supply driver which sequentially selects said plurality of voltage supply lines in synchronism with the sequential selection of said plurality of selection scan lines by the selection scan driver.

6. An apparatus according to claim 5, wherein each of said pixel circuits comprises:
a first transistor having a gate connected to the selection scan line, and a drain and source one of which is connected to the current line;
a second transistor having a gate connected to the selection scan line, and a drain and source one of which is connected to the voltage supply line;
a driving transistor having a gate connected to the other of the drain and source of the second transistor, and a drain and source one of which is connected to the voltage supply line, and the other of which is connected to the other of the drain and source of the first transistor; and
a capacitor which stores a gate-to-one of source and drain voltage of the driving transistor by holding the voltage.

7. An apparatus according to claim 6, which further comprises a plurality of light-emitting elements which are arranged at intersections of said plurality of selection scan lines and said plurality of current lines, emit light at luminance corresponding to a current value of a driving current, and each have two electrodes one of which is connected to a corresponding one of said plurality pixel circuits, and
in which the other electrode of the light-emitting element is connected to the other of the drain and source of the driving transistor.

8. An apparatus according to claim 7,
wherein in the selection period, the first transistor supplies the designating current from the voltage supply line to the current line via the drain-to-source path of the driving transistor, the driving transistor converts the current value of the designating current into a level of a gate-to-one of source and drain voltage, and the capacitor stores the level of the converted voltage, and
after the selection period, the driving transistor supplies, to the light-emitting element, a driving current having a current value corresponding to the level of the gate-to-one of source and drain voltage stored by the capacitor.

9. An apparatus according to claim 7, wherein the voltage applied to the voltage supply line by the voltage supply driver in the selection period is set not higher than a voltage of the other electrode of the light-emitting element, and the voltage applied to the voltage supply line by the voltage supply driver after the selection period is set higher than the voltage of the other electrode of the light-emitting element.

10. A display device comprising:
a plurality of selection scan lines;
a plurality of current lines;
a plurality of light-emitting elements which are arranged at intersections of said plurality of selection scan lines and said plurality of current lines, and emit light at luminance corresponding to a current value of a driving current;
a selection scan driver which sequentially selects said plurality of selection scan lines in each selection period;
a data driving circuit which applies a reset voltage to said plurality of current lines in a first part of the selection period, and supplies a designating current having a current value corresponding to an image signal to said plurality of current lines in a second part of the selection period after applying the reset voltage in the selection period; and
a plurality of pixel circuits which are connected to said plurality of selection scan lines and said plurality of current lines, and electrically connect said plurality of current lines and said plurality of light-emitting elements to each other in the selection period;
wherein in the selection period, each of said plurality of pixel circuits loads the designating current which flows through said plurality of current lines, and stores a level of a voltage converted in accordance with the current value of the designating current, and after the selection period, each of said plurality of pixel circuits shuts off the designating current which flows through said plurality of current lines, and supplies a driving current corresponding to the level of the voltage converted in accordance with the designating current.

11. A display panel driving method comprising:
sequentially selecting a plurality of selection scan lines of a display panel comprising a plurality of pixel circuits connected to the plurality of selection scan lines and a plurality of current lines, and a plurality of light-emitting elements which are arranged at intersections of the plurality of selection scan lines and the plurality of current lines, wherein each of the light-emitting elements emits light at luminance corresponding to a current value of a current flowing the current line;
applying a reset voltage to the plurality of current lines in an initial part of a period in which each of the plurality of selection scan lines is selected;
after applying the reset voltage, supplying designating currents having current value corresponding to an image signal to the plurality of current lines, and storing, in the plurality of pixel circuits, the current value of the designating currents flowing through the plurality of current lines; and
after supplying the designating currents, allowing the plurality of pixel circuits to supply, to the plurality of light-emitting elements, driving currents having current value corresponding to the stored current value of the designating currents.