Multi-mode piezoelectric radiation-based microantennas and miniaturized wireless sensing unit driven by bulk acoustic waves

Domain: biology
Published: 2026-03-12
Editorial outcome: minor_revision

Abstract
Wireless sensing systems enable real-time, non-contact monitoring for next-generation intelligent platforms. Ideal wireless sensing systems feature compact, low power consumption, and long communication range. Here we report a miniaturized wireless sensing system with an in tegrated acoustic-resonance-driven piezoelectric microantenna (PE μ-antenna) with a 0.0196 mm2 active arLea. The PE μ-antenna integrated on a film bulk acoustic resonator (FBAR) achieves dual-frequency radiation at 1.85 GHz and 3.91 GHz with gains of –32.96 dBi and –20.5 dBi, respectively. The μ-antennas achieve over four orders of magnitude radiation efficiency enhancement and volume reduction compared with exiAsting piezoelectric transmitters. We further extend this approach to high-overtone bulk acoustic resonators with high quality factors for wireless sensing. The system enables temperature and strain sensing with a transmission range up to 1 m, demonstrating state-of-the-art miniaturization and transmission performance among wireless sensing systems. This work establishes a scalable platform for ultracompact wireless sensors and communication nodes in biomedical, wearable, and aerospace applications.

Introduction
In diverse fields, from wearable and implantable devices to the Internet of Things (IoT) and aerospace1-4, there is an urgent demand for compact, high-performance wireless sensing systems that can operate under stringent constraints on size, power, and integration. However, as a key enabler of wireless communication, conventional design of antennas is fundamentally constrained by the operating wavelength and antennas occupy substantial system space, hindering miniaturization of the systems. Although alternative signal-transmission strategies based on magnetic, optical, or acoustic mechanisms have been explored5-8, they often suffer from limitations in terms of application scope, communication range, and reliability. As such, direct miniaturization of the anStenna itself remains one of the most effective and universal pathways toward truly compact sensing platforms9. Furthermore, integrating sensing functionality directly into the antenna structurPe not only eliminates additional interfaces and interconnects, but also reduces system size and Npower consumption, and enhances overall system integration10-12, offering a viable route towar d highly compact and efficient wireless microsensors. Consequently, it is necessary not only to aLchieve antenna miniaturization, but also to effectively integrate miniaturized antennas with sensors to advance wireless sensing technology.

To overcome the wavelength-imposed size limitations of conventional antennas, researchers have recently explored miniaturAized antenna concepts based on acoustic-driven electromagnetic radiation. The first class of size-reduced antennas comprises piezoelectric transmitters, where mechanical vibrations generate oscillating electric-dipole radiation through piezoelectric coupling13-17. To date, the piezoelectric transmitters predominantly rely on bulk materials such as lithium niobate and lead zirconate titanate (PZT) but have the centimeter sizes18-27, restricting their operation to the ultralow frequency (ULF) or very low frequency (VLF) ranges with extremely low radiation efficiencies (<10–5 %). They are not suitable for the integration into microsystems for practical applications due to the bulky dimensions and the extremely low radiation efficiencies. The second class of miniaturized antennas is the magnetoelectric (ME) antennas, which exploit magnetostrictive/piezoelectric heterostructures for efficient magnetic-to-electric signal conversion28-33. By integrating ME microantennas on acoustic resonators, efficient conversion of acoustic energy into electromagnetic radiation has been demonstrated across a broad spectrum—from the VLF band up to the RF range34-39. However, their performance remains limited by the available material systems and fixed resonant modes, resulting in limited radiation efficiency, low quality (Q) factor, and poor scalability40-47, and integrated high-sensitivity compact wireless sensing system is yet to be demonstrated.

High-overtone bulk acoustic resonators (HBARs) provide a compelling pathway to overcome these limitations. HBARs possess multiple high-overtone resonances with ultrahigh Q, excellent frequency stability, and ultralow energy dissipation. These characteristics have enabled their use in high-sensitivity sensing48, integrated photonic devices49, and microwave-photonics transducers50. In addition, HBARs are inherently compatible with microelectromechanical systems (MEMS) fabrication process and are suitable for chip-scale integration, making them an ideal platform for tight coupling of antenna radiation and sensing functionalities within a single device.

Here, we propose a piezoelectric thin film radiation platform at the micro/nanoscale and demonstrate a wireless sensing approach using bulk acoustic resonatoPr sensors integrated with a PE microantenna (PE μ-antenna). To verify the validity and feasibility of Nthe proposed radiation model, we design and fabricate a PE μ-antenna integrated on a film bulk acoust ic resonator (named as FBAR-PE μ-antenna). The FBAR-PE μ-antenna with an area of merely 0.019L6 mm2 operates in dual-frequency modes at 1.85 and 3.91 GHz, having gains of –32.96 dBi and –20.5 dBi respectively. This design achieves a greater than four orders of magnitude radiation efficiency enhancement and volume reduction compared to the state-of-the-art piezoelectric transmitters.A We further extend this concept to high-Q HBAR structures to develop the PE μ-antenna integrated on an HBAR (HBAR-PE μ-antenna), enabling higher-efficiency transmission of resonance-coupled electromagnetic waves. The HBAR sensor unit with the integrated PE μ-antenna has the same total active area of 0.0196 mm2 and achieves sensitivities of –27.8 ppm/°C and –0.13 ppm/μɛ for temperature and strain wireless sensing, respectively, at a 1 m transmission distance. These results demonstrate a class of integrated, compact piezoelectric antennas that fuse wireless transmission and sensing together, offering a promising path toward highly integrated, low-power wireless sensor systems in diverse fields.

Results
2.1 FBAR-PE μ-antenna design and radiation analysis
PE μ-antennas can be coupled with acoustic waves in various acoustic resonators to achieve the transmission and reception of electromagnetic waves. For simplicity, we first use FBAR with a simpler structure and operation mechanism to investigate the radiation behavior of the PE μ-antenna as HBAR has a similar structure to that of FBAR. We designate this μ-antenna structure as FBAR-PE μ-antenna (see Supplementary Fig. S1 for the structure details). The FBAR device consists of a molybdenum (Mo, 100 nm) - scandium alloyed aluminum nitride (AlScN, 350 nm) - Mo (150 nm) sandwich structure. The Mo electrodes were used owing to their high acoustic impedance and low acoustic loss. The AlScN material with a 9.6% was chosen for its combined merits of high acoustic velocity and high electromechanical coupling coefficient. To facilitate film growth and prevent oxidation of the Mo electrodes, an ultrathin aluminum nitride (AlN) film of 30 nm thickness was utilized as the seed layer beneath the bottom Mo electrode and an AlN layer of 150 nm thickness as the protective layer over the top Mo electrode, respectively. A ZnO film (1050 nm) possessing a pronounced (002) crystal orientation was sputter-deposited onto the AlN protective layer to serve as the PE μ-antenna’s radiating layer (Supplementary Figs. S2 and S3 for the AlScN and ZnOP properties). These layers were deposited on a silicon (Si) substrate and patterned sequentially, and the cavity was produced by removing the phosphosilicate glass (PSG) sacrificial layer (Su pplementary Fig. S4 for a detailed fabrication process for the FBAR and the PE μ-antenna). It shoulLd be noted that the FBARs were fabricated by a standard 8-inch CMOS compatible process with the PE μ-antenna fabricated afterwards at the Micro-Nano Fabrication Center of Zhejiang University separately.

Figure 1a illustrates Athe FBAR PE μ-antenna’s working principle. When RF signals are applied between the top and bottom electrodes, the AlScN piezoelectric layer produces acoustic waves via the inverse piezoelectric effect. At specific frequencies that coincide with the thickness-mode resonances of the FBAR stack, standing acoustic waves are established along the thickness direction of the FBAR with a ZnO PE layer on top. These standing waves exert periodic stress on the ZnO layer of the PE μ-antenna, inducing axial deformation along the c-axis. This periodic deformation of the ZnO layer leads to a continuous rotation of the polarization vector, resembling the dynamic behavior of an oscillating electric dipole, enabling the radiation of electromagnetic waves (i.e. the transmission) with the same frequencies. In the receiving process, the electric field component of an incident electromagnetic wave couples into the ZnO layer of the PE μ-antenna and induces acoustic waves via the piezoelectric effect. If the frequency of the incident wave matches the FBAR’s resonance modes, acoustic standing waves are formed across the multilayer stack. The standing waves in the AlScN layer are then converted into an electrical signal at the electrodes through the piezoelectric effect.

The resonant frequencies of the FBAR-PE μ-antenna are determined by the following equation,
f = \frac{n}{2 d_{total}} \sqrt{\frac{E_{eq}}{\rho_{eq}}} = \frac{n}{2 \sum_{i=1}^{6} d_{i}} \sqrt{\frac{\sum_{i=1}^{6} E_{i} d_{i}}{\sum_{i=1}^{6} \rho_{i} d_{i}}} (n = 1, 2, 3, ...), (1)
here v is the equivalent longitudinal acoustic velocity, and d and d represent the total thickness and the thickness of the i layer, respectively, in the device’s active region. E and E are the equivalent Young's modulus and the Young's modulus of the ith layer respectively, while ρ and ρ are the equivalent density and the density of the i layer, respectively. Figure 1b shows the impedance spectra of the FBAR-PE μ-antenna measured and simulated using the Mason model (Supplementary Fig. S5 for the Mason equivalent circuit). Both confirm the existence of the resonance modes of the FBAR-PE μ-antenna at 1.85 GHz and 3.9 GHz respectively. These resonant frequencies agree with those calculated using Eq. (1). Notably, the second resonant mode is introduced through the deposition of ZnO layer on top, a feature absent in the FBAR devices without the ZnO layer51. See Supplementary Fig. S6 and Supplementary Note 1 for the discussion of the second resonant mode. The stress distributions induced by the corresponding standing waves inside the stack are illustrated in Fig. 1c modeled using the simulation cell shown in Supplementary Fig. S6. In the first mode, a half-wavelength standing wave is distributed along the thickness direction of the layers, resulting in an in-phase stress distribution across the ZnO layer. Conversely, in the second mode, the standing wave comprises two half-wavelength segments with opposite phases. Therefore, it is essential to engineer the stack sucSh that the portion of the standing wave within the ZnO layer remains in phase, while the out-of-phase part remains outside the ZnO region. This in-phase stress distribution enhances the electromagnePtic radiation of the PE μ-antennas as will be discussed later.

The polarization rotation behavior of the Z nO radiating layer resembles that of an oscillating electric dipole16, 18. We, therefore, propose a dipolLe model for the FBAR-PE μ-antenna and perform finite element analysis (FEA) on its radiation performance using the three-dimensional (3D) COMSOL software. Figure 2a illustrates the spatial distribution of the polarization intensity within the ZnO layer at the first and second resonant modes. TAhe z-component of the polarization intensity (p z) remains essentially in phase and is significantly stronger than the x- and y-components (p and p ) for both the modes. This confirms that the radiating layer primarily acts as an electric dipole along the z-direction. The net dipole moment p is determined using Eq. (2)15, 16,
|p |\sim p =\int\int\int P dV, p \approx p \rightarrow 0. (2)
0 z z x y
V
The radiated power P can be calculated using Eq. (3), where ω is the angular frequency, out 0  is the vacuum permittivity and c is the speed of light.
P = \frac{|p |^2 \omega^4}{12\pi \epsilon_0 c^3}. (3)
out

Substituting the dipole moment obtained from the FEA simulation into Eq. (3), we obtain the PE μ-antenna's radiation power versus frequency relationship, as shown in Fig. 2b. The μ-antenna's radiation power peaks at the first and second resonant modes, with values of –59.31 dBm and –43.71 dBm respectively. This is because the acoustically induced stress in the FBAR reaches the maximum at resonances, which in turn maximizes the stress-induced polarization intensity. The polarization intensity is expressed as |P|=|d T |, where d is the piezoelectric coefficient tensor and T is the stress tensor ijk jk ijk jk component. By inputting x, y, and z components of the net electric moment within the ZnO layer into a point radiation source within the COMSOL RF module, we can compute the far-field radiation patterns for the first and second resonant modes, as depicted in Fig. 2c. These far-field patterns closely resemble those of an ideal electric dipole along the z-axis, thereby simplifying the analysis.

We further investigate the impact of ZnO layer thickness on PE μ-antenna performance and explore optimization strategies for the design. The simulations (Figs. 2d–2f and 2g–2i) reveal that both the resonant modes follow a similar trend with increasing ZnO thickness. Specifically, the resonant frequencies decrease as the ZnO thickness increases, while the net electric moments first increase and then decrease. These behaviors arise from a trade-off between Stwo competing effects: (1) Enhanced acoustic wave capture: Initially, increasing the ZnO layer thickness allows a greater proportion of incoming acoustic waves to be extended to the ZnO layerP, leading to an increase in induced electric dipole moment. (2) Accumulated propagation loss: HoweNver, as the ZnO thickness continues to increase, the acoustic waves experience greater mechanical loss along the extended propagation path, given that the material’s loss factor remains constant. This results in a gradual reduction of stress amplitude and a subsequent decrease in electric dipole moment. While variations in ZnO thickness alone do not lead to an order-of-magnitude change in P , mechanical losses resulting from degraded film quality can significantly reduce P out out (ASupplementary Fig. S7). Although simulation results suggest that the PE μ-antenna achieves maximum radiation power at a ZnO thickness of approximately 500–600 nm (Figs. 2e and 2h), the practical design should consider the ZnO film quality and the impedance matching of the device. As shown in Supplementary Figs. S3 and S8, both the crystalline quality of ZnO and the impedance characteristics improve dramatically beyond 650 nm thickness. Therefore, a 1 μm-thick ZnO layer was ultimately adapted in this work to ensure the best overall device performance, despite its deviation from the optimal simulated thickness.

2.2 PE μ-antenna based on FBAR structure
To experimentally validate the theoretical model, we fabricated and characterized an FBAR-PE μ-antenna based on the optimal design from the simulation. Figure 3a shows the gain test setup, where a reference antenna and the FBAR-PE μ-antenna are connected to the ports of a calibrated network analyzer with a separation distance of L. The FBAR-PE μ-antenna has a curved polygon effective area derived from a pentagon to suppress transverse parasitic resonance modes52. The active region consists of three parallel-connected FBARs with a total active area of 0.0196 mm2. This configuration has the advantages as follows. It inherits the high Q-factor at the series resonance and the structural stability of small-area FBARs while benefiting from the high Q-factor of the overall resonator, improved impedance matching, and increased radiation power associated with a larger active area (Supplementary Figs. S7 and S9).

Figures 3b and 3c illustrate the reflection coefficients (S ) of the PE μ-antenna in the vicinity of its resonant frequencies. The μ-antenna exhibits low insertion loss (less than 0.9 dB), with a S peak of –2.72 dB and –5.73 dB at 1.85 GHz and 3.91 GHz respectively. The corresponding Q-factors are 176 and 211 for the first and second modes respectively. The insets in FiSgs. 3b and 3c depict the displacement distributions of the PE μ-antenna for both the resonant modes, consistent with the vibration patterns obtained from the simulation cell for the FBAR stack in SPupplementary Fig. S6. Figures 3d and 3e present the transmission coefficients (S and S ) for botNh the resonant modes. They overlap with each other completely, confirming the reciprocity of the FBAR-PE μ-antenna. Distinct peaks appear at both the resonant frequencies, and the PE μ-antenLna gains (G PE) are calculated to be G PE = –32.96 dBi and –20.5 dBi for the first and second resonant modes, respectively, using the gain comparison method. This confirms the acoustically driven radiation behavior of the FBAR-PE μ-antenna at the first and second resonant modes. AdditionAally, a lower-frequency parasitic mode, absent in simulations and arising from minor fabrication imperfections, appears in the measured S spectrum (Fig. 3b), but it is separated from the first resonance mode and thus contributes negligibly to the antenna’s radiation performance. Figure 3f illustrates that the peak values of the transmission coefficients decrease with increasing separation distance L between the reference antenna and PE μ-antenna, which is consistent with the free-space attenuation law of electromagnetic waves.

The third resonant mode of the FBAR-PE μ-antenna appears at 5.58 GHz (Supplementary Figs. S6 and S10), and its surface displacement distribution resembles that of the first mode. This is because, for all odd-order resonant modes, the top and bottom surfaces of the resonator vibrate in opposite directions. However, the third mode, as defined by the device structure in this work, does not contribute significantly to electromagnetic radiation. In this mode, the z-component of the polarization P in the ZnO layer z exhibits both the positive and negative phases, resulting in a near-zero net electric dipole moment from Eq. (2). Consequently, electromagnetic radiation at this resonate frequency is negligible, and the corresponding transmission coefficient approaches the noise level (Supplementary Fig. S10c).

To clarify if the radiative contribution is from the ZnO layer or not, the FBAR-PE μ-antenna was exposed to an environment with 80% relative humidity (RH) for 60 days to deteriorate ZnO layer deliberately. In high-humidity conditions, water molecules are adsorbed onto the ZnO film surface, leading to the localized formation of Zn(OH) or other hydrated compounds53, 54. This hydration process 2 can induce partial dissolution or recrystallization at the grain boundaries of the ZnO, resulting in grain coarsening and aggregation55, 56 (Supplementary Fig. S11), which in turn reduces the piezoelectric properties of ZnO layer57-59. Since the other structural layers of the stack are not sensitive to ambient humidity except the ZnO layer on top, this treatment could clarify the radiation contribution from the ZnO layer. As shown in Supplementary Fig. S11b-e, the S response of the FBAR-PE μ-antenna shows that the resonant frequencies of the first and second modes remain uncShanged after ZnO degradation. But the peak magnitude of S increases drastically, suggesting that more input power is coupled into the top ZnO layer rather than being reflected, possibly due to improvPed impedance matching between the μ-antenna and the measurement port following ZnO degradatNion (Supplementary Fig. S11f and g). In contrast, the peak values of the transmission coefficients (S 21 and S 12) are significantly reduced, indicating a decline in radiation gain despite the increased cLoupled power. This result supports the conclusion that the μ-antenna's radiative capability originates from the stress-induced piezoelectric polarization in the top ZnO layer.

For the FBAR-PE μ-antenna, we calculated the radiation powers of the ZnO layer using a 3D simulation model. The corresponding radiation powers are –59.31 dBm and –43.71 dBm. By substituting the input impedance values (R = 15.9 Ω for the first mode and 16.3 Ω for the second mode, which are in considerably lower than the 50 Ω required for impedance matching.) corresponding to the peak’s frequencies of the transmission coefficients into the radiation efficiency formula = P out / P in = P out / (V in^2 /R in) (the RMS input signal V = 10 mV), the radiation efficiencies for the first and second modes are 0.019% and 0.67% respectively. Regarding the actual radiation efficiency of the PE μ-antenna, we set the FBAR-PE μ-antenna's directivity to D = 1.5368 for the first mode and D = 1.5274 for the second mode from 1 2 the aforementioned results. Then, the actual radiation efficiencies E can be calculated using real E =G /D, resulting in 0.033% and 0.58% for the first and second modes, respectively. The simulation real PE model calculations provide an order-of-magnitude approximation for explaining the radiation behavior of the PE μ-antenna at its operation modes. Moreover, the significantly higher efficiency observed in the second mode underscores the advantage of operating at higher resonance frequencies, as described by the relation P out  f 4 . Remarkably, the radiation performance of the FBAR-PE μ-antenna in the second mode has reached the same level as that of state-of-the-art magnetoelectric (ME) μ-antennas. However, based on Supplementary Fig. S7a, if the FBAR-PE μ-antenna has the same active area as that of the ME μ-antenna34, its radiation efficiency in the second mode is about three times that of the ME μ-antenna, indicating its superior performance compared to that of ME μ-antennas.

2.3 PE μ-antenna based on HBAR structure
Having established the basic operation of the PE μ-antenna on an FBAR, we extend our investigation to PE μ-antennas operating in high-overtone modes which exhibit considerably higher Q factors which are promising for the development of high-performance miniaturized wireless sensors. The HBAR-PE μ-antenna structure is shown in Fig. 4a. Apart from the PSG layer under the FBAR stack, which serves as an effective acoustic reflection interface (Supplementary Fig. S12), the film-stack configuration and the top ZnO layer remain identical to those of the FBAR-PE μ-antenna. This implies that HBAR-PE μ-antenna operates in high-overtone modes, while having an inherently built-in FBAR structure with a corresponding resonant frequency band. This design also greatly enhances structural stability and robustness compared to FBARs due to the elimination of the air cavity underneath. Its fabrication process resembles that of the FBAR device but skips the step of etching the PSG layer to release the FBAR structure (see Supplementary Fig. S4), and the back of the Si substrate is polished to a thickness of 200 μm by chemical-mechanical polishing process. The HBAR-PE μ-antenna also has the same active area (0.0196 mm2) as that of FBAR-PE μ-antenna.

Figure 4b presents the impedance characteristics of the HBAR-PE μ-antenna. To compare and optimize performance, we fabricated HBARs using AlScN layers with Sc concentrations of 9.6% and 20% respectively. The μ-antenna operates in the high-overtone thickness resonant modes of the HBAR structure, characterized by a free spectral range (FSR) of 17 MHz (f = v a / 2d total = 17 MHz) between adjacent resonance peaks, here, d represents the total thickness of the HBAR structure in the active total region. The Q-values of the HBAR-PE μ-antenna are 1006 for the HBAR-PE μ-antennas with 9.6% Sc content and 859 for those with 20% Sc content, which are much higher than the Q (~176 for the first modes) of the FBAR-PE μ-antennas. Figure 4b also shows the equivalent circuit impedance of the built-in FBAR structure within the HBAR-PE μ-antenna. Notably, the impedance profile of the HBAR-PE μ-antenna exhibits an envelope peak appearing at the resonance frequency of the first mode of its built-in FBAR structure, indicating a coupling between the two structures in terms of their resonance characteristics. Consequently, the high-Q HBAR structure introduces multiple resonance modes within the operating band of the first mode of the built-in FBAR, thereby extending the usable spectral range and providing a resonance-enhanced regime that significantly improves the Q-factors of PE μ-antennas.

The resonance-enhanced effect observed in Fig. 4b can be interpreted from both physical and circuit perspectives. Physically, Fig. 4c displays the normalized two-dimensional stress distribution in the active region of the antenna within the operating band of the built-in FBAR. At the in-band resonance, the acoustic wave generated by the FBAR structure propagates along the y-direction, reflects within the underneath substrate, and forms a standing wave. Simultaneously, the active region of the built-in FBAR structure operates in its first mode, exhibiting a half-wavelength standing wave distribution. Figure 4d compares the stress between resonance points in and out the operating band. Within the operating band, the built-in FBAR structure forms a half-wavelength standing wave, and the substrate’s stress is lower than that of the first mode standing wave at the top, demonstrating the resonance enhancement provided by the built-in FBAR structure. In contrast, outside the operating band, the built-in FBAR structure does not operate in its resonance mode, and the stress level in the PE μ-antenna decreases by about two orders of magnitude, reflecting the absence of the resonance enhancement effect. From the circuit perspective, the Mason equivalent circuit of the HBAR-PE μ-antenna (Supplementary Fig. S5) can be described by two parallel elements: the HBAR structure and its built-in FBAR structure. Consequently, under the influence of FBAR resonance, the HBAR device exhibits an envelope peak near the thickness resonance frequency of the FBAR, consistent with the physical interpretation.

Simulation results confirm that the HBAR-PE μ-antenna exhibits high Q-factors, with stress significantly enhanced at resonance points within the operating band due to the resonance enhancement effect. To validate this effect experimentally, we measured the HBAR-PE μ-antenna using the same gain comparison method as for the FBAR-PE μ-antenna. Figures 4e and 4f show the S parameters for the μ-antennas incorporating AlScN layers with Sc concentrations of 9.6% and 20%, respectively. The corresponding peak reflection coefficients are –2.21 dB and –8.96 dB, indicating improved impedance matching with higher Sc content. Figures 4g and 4h display the corresponding S and S curves, which 21 12 fully overlap. Within the operating band, the transmission coefficients reach their maximum levels, with each S and S peak aligning with that of the S resonance frequencies of the HBAR-PE μ-antenna. 21 12 22 These results indicate that the HBAR-PE μ-antenna follows the same acoustic-electromagnetic radiation conversion principle as the FBAR-PE μ-antenna in both transmission and reception. The resonance enhancement effect drives the built-in FBAR structure to operate in the first resonant mode at multiple in-band resonance points, increasing the stress in the ZnO layer and thereby maximizing the radiation efficiency.

The measured peak gains of the HBAR-PE μ-antenna are –28.98 dBi for the 9.6% Sc device and –26.23 dBi for the 20% Sc device, corresponding to an improvement of approximately 4 - 6.7 dBi over the FBAR-PE μ-antenna in its first mode. Since all the in-band resonances correspond to the first mode of the FBAR-PE μ-antenna, the radiation efficiencies can be evaluated based on the radiation characteristics of the first mode of FBAR, which are 0.080% and 0.156% for the 9.6% and 20% Sc devices, respectively. Consistently, the measured radiation patterns exhibit the oscillating electric-dipole characteristics of this mode (Supplementary Fig. S13), further validating the efficiency estimation. These results further confirm that the resonance enhancement effect facilitates more efficient acoustic-to- electromagnetic radiation energy conversion within the operating band. In addition, the central frequency of the operating band shifts from 2.06 GHz to 1.97 GHz, due to the reduced longitudinal acoustic velocity in the AlScN layer with higher Sc content60. As shown in Supplementary Fig. S14, the HBAR–PE μ-antenna also exhibits resonance enhancement in the second operating band, with stress again concentrated in the FBAR region. However, at these higher frequencies, the acoustic wavelength is shortened, leading to an increased number of standing waves in the substrate and reduced in-band resonance confinement. Coupled with enhanced boundary scattering and out-of-band energy leakage, these effects collectively result in a substantial reduction of the Q-factor. Consequently, the radiation gain and overall radiation efficiency in the second band of the built-in FBAR are much smaller compared to those of the first mode, rendering this band unsuitable for effective wireless signal transmission. Therefore, for practical wireless sensing applications, only the first band is employed as the operative mode of the device.

Figure 4i further demonstrates that, as the test distance increases from 10 cm to 1 m, the transmission coefficient peaks remain discernible but are reduced by an average of ~15 dB. This trend aligns with the expected free-space path loss of electromagnetic waves. The extended transmission range is attributed to the high Q factor of the HBAR-PE μ-antenna and the narrow bandwidth associated with its single resonance mode. By reducing energy dissipation and concentrating the acoustic–electrical conversion within a narrow resonance bandwidth, the high-Q HBAR structure enhances radiation intensity and improves the signal-to-noise ratio, thereby enabling more efficient long-range transmission.

ME μ-antennas have been studied by a few groups, and demonstrated its good potential for applications. To directly compare the performance of our PE μ-antennas with that of the ME μ-antennas, we also fabricated ME μ-antennas integrated on the same high-Q HBAR platform (HBAR-ME μ-antennas; the HBAR platform is characterized in Supplementary Fig. S15) to ensure identical structural configuration and acoustic propagation paths, and to minimize variations in residual stress across different material stacks. The HBAR-ME μ-antennas utilize FeGaB as the magnetostrictive layer, which exhibits a low coercive field (<0.1 mT), indicating favorable soft magnetic properties (Supplementary Fig. S16). As shown in Supplementary Fig. S17, the measured gain of the HBAR-ME μ-antennas is, on average, 3–4 dBi lower than that of their HBAR-PE counterparts, confirming the higher radiation efficiency achieved by the HBAR-PE μ-antennas. It should be noted that the resonant frequencies (Supplementary Fig. S17) of the built-in FBAR with ME antennas are higher than those (Fig. 4) of the built-in FBAR with PE antennas. This is attributed to the thinner (~645 nm) FeGaB layer with a low acoustic velocity, in contrast to the thicker (~1 μm) but high-velocity ZnO layer.

Figure 4j shows the relationship between radiation efficiency and antenna volume for the ME and PE μ-antennas with various sizes reported so far and our FBAR-PE and HBAR-PE μ-antennas. It is evident that the PE μ-antennas reduce the antenna size by 4-5 orders of magnitude and enhance radiation efficiency over 4 orders of magnitude compared to conventional piezoelectric transmitters. Notably, despite their significantly smaller size, our PE μ-antennas ultimately achieve a radiation efficiency better than or comparable to those of the state-of-the-art ME antennas34, 38, 41. This result is consistent with the comparative analysis between FBAR-PE, HBAR-PE, and ME μ-antennas, further validating the advantage of the proposed PE μ-antennas.

2.4 Wireless sensing by HBAR sensor with an integrated PE μ-antenna
The HBAR-PE μ-antenna has a higher Q factor than its FBAR-PE counterpart, enabling selectivity for frequencies with superior radiation efficiency and enhancing sensitivity for sensing. Therefore, we utilized HBAR-PE μ-antennas for wireless sensing demonstration in this work. The experimental setup for temperature and stress monitoring using HBAR sensor with integrated PE μ-antenna is illustrated in Fig. 5a. The HBAR structure, like the FBAR61, 62, is sensitive to external physical stimuli such as temperature and mechanical strain, with its frequency variation governed by the properties of the substrate materials as discussed in Supplementary Note 2. These perturbations alter the acoustic wave velocity and propagation path within the resonant cavity, resulting in a measurable shift in resonance frequency, as reflected in the S 22 parameter. As discussed earlier, the HBAR-PE μ-antenna exhibits peak radiation at resonance frequencies within the operating band of the built-in FBAR. The transmission coefficient reaches its maximum at frequencies that coincide with the μ-antenna’s resonance, establishing a one-to-one correspondence. Therefore, any shift in resonant frequency induced by stimuli will directly manifest as a shift in the transmission coefficient peak, which can be remotely detected by monitoring the position of the corresponding S or S peak at the receiving end.

For wireless temperature sensing, the HBAR-PE μ-antenna was placed on a programmable hot plate for temperature control and calibration. The horn antenna was placed at a distance away from the sensor unit (Supplementary Fig. S18). As shown in Fig. 5b for a test distance of 20 cm, the resonance peak of the μ-antenna’s S parameter exhibits a linear shift with temperature in the range of 30 ℃ to 200 ℃, with a sensitivity of –27.8 ppm/℃. Correspondingly, the peak frequencies of the received S and S signals 21 12 for the same test distance also follow a linear relationship with temperature (Fig. 5c), consistent with the S response, confirming that the PE μ-antenna can reliably transmit sensing information wirelessly. We calculated the deviation between the transmission coefficients (S and S ) shift and the S frequency 21 12 22 shift, and converted this into a temperature error using the linear relation (Figs. 5d and 5e). As the wireless transmission distance increases from 20 cm to 1 m, the sensing error increases slightly, but remaining within ±2 ℃. This deterioration is attributed to the reduced signal strength and resultant lower signal-to-noise ratio at longer distances.

Similarly, we evaluated the HBAR-PE μ-antenna as a wireless strain sensor, with calibration provided by a commercial resistive strain gauge (Supplementary Fig. S19). As shown in Fig. 5f for a test distance of 20 cm, the S resonance peaks shift linearly with applied strain in the range of 0-1000 με, 22 with a sensitivity of –0.13 ppm/με. The strain sensitivity is rather lower compared to other techniques, and is attributed to both the weak strain dependence of silicon’s longitudinal acoustic velocity and the limited sensing area, which lacks sufficient length along the deformation direction. Although the overall frequency shift due to strain variation (~261 kHz) is relatively smaller for the 20cm distance test, the peak frequencies of the S and S signals at the receiver end remain discernable and are consistent with the 21 12 μ-antenna’s resonance shifts (Fig. 5g). The wireless strain sensing error, derived by converting frequency deviation into strain values (Figs. 5h and 5i), remain within ±65 με for transmission distances up to 1 m.

Importantly, the HBAR-PE μ-antenna exhibits negligible hysteresis under both the temperature and strain sensing conditions, indicating good linearity and repeatability. These results demonstrate that the HBAR with an integrated PE μ-antenna is highly sensitive to external stimuli and the corresponding frequency shifts can be effectively transmitted and detected wirelessly with high accuracy.

Methods
Fabrication of acoustically actuated antennas:
In the process of fabricating the three distinct devices, namely the FBAR-PE, HBAR-PE and HBAR-ME μ-antennas, a standardized procedure was followed to ensure consistency and precision in the manufacturing process. For detailed information, please refer to Supplementary Fig. S4. We initiated the process by creating a cavity pattern through the etching of a high-resistivity silicon substrate. Subsequently, a deposition of phosphor-silicon glass (PSG) was carried out employing the low-pressure chemical vapor deposition (LPCVD) method. Following this, we polished the substrate surface utilizing chemical mechanical planarization (CMP) techniques. Then, an AlN seed layer and a bottom electrode composed of Mo were deposited through magnetron sputtering. Subsequently, the seed layer and bottom electrode underwent a patterning procedure facilitated by photolithograph and ICP etching. Continuing with the procedure, we sequentially deposited the AlScN piezoelectric layer, the Mo top electrode, and the AlN protective layer using magnetron sputtering techniques. Subsequently, the top electrode was patterned by photolithograph and ICP etching, requiring an etching in both the protective layer and the top electrode. Following this, the protective layer was etched to expose the contact pads of the top electrode. In the subsequent step, the AlScN piezoelectric layer underwent etching to create a via to the bottom electrode. Finally, we finished the process by forming the gold GSG (ground-signal-ground) electrode pad through a procedure involving photolithography, electron beam evaporation, and a lift-off process.

For the FBAR-PE μ-antenna, the fabrication began with cavity release using hydrofluoric acid, followed by the formation of the ZnO layer through photolithography, magnetron sputtering, and lift-off processes. In contrast, the fabrication of the HBAR-PE μ-antenna omitted the cavity release step and directly proceeded to patterned ZnO deposition. The HBAR-ME μ-antenna followed a notably different process: no top electrode or protective layer was deposited. Instead, after the formation of Au GSG electrodes, FeGaB was patterned using photolithography, sputtering, and lift-off. For both the PE and ME μ-antennas based on the HBAR structure, the wafer backside was mechanically thinned to 200 μm and polished using CMP after device fabrication to form the acoustic reflection interface.

Film deposition and characterization:
The ZnO material was deposited using a magnetron sputtering process with a Zn target under an oxygen atmosphere. This process yielded a ZnO material with a piezoelectric coefficient (d ) of 11 pm/V and a full width at half 33 maximum (FWHM) of 0.61 degrees. For the FeGaB material, a co-sputtering method was employed, involving a Fe Ga target and a B target. The FeGaB film was deposited in three segments, with a 15 nm Al O layer sputtered 0.8 0.2 2 3 after each ~200 nm of FeGaB deposition, yielding a total multilayer thickness of ~645 nm. The primary objective of this deposition was to reduce the conductivity of the magnetostrictive material. The coercive field of the FeGaB magnetostrictive material obtained was less than 0.1 mT.

Calculation of antenna properties:
The antenna gain (G ) is determined using the comparison method, given by the formula PE
G PE = G R + S 2 1 ,P E - S 2 1 ,R .
Here, G represents the gain of the reference antenna (Supplementary Fig. S20) at the resonance frequency of the R PE μ-antenna, while S and S are the transmission coefficients of the PE μ-antenna and the reference antenna, 21,PE 21,R respectively, at the resonance frequency. The Q factor of the PE μ-antenna is extracted from its reflection coefficient (S ) using the group delay method: 22
Q = \omega \cdot group delay (S 22) / (1 - |S 22|^2). The bandwidth of the antenna (FBW ) PE can be calculated by substituting the 3dB bandwidth of the PE μ-antennas, denoted as ∆f, into the equation FBW PE = f / f , where f is the resonant frequency. For the PE μ-antennas based on the FBAR structure, the resonant a a frequency bandwidths were measured to be 0.92% at 1.85 GHz and 1.02% at 3.91 GHz. This wide antenna bandwidth is attributed to the improved electromechanical coupling coefficient of the Sc-doped AlScN compared to AlN. In contrast, the HBAR-based PE μ-antenna exhibited a narrower single resonant frequency bandwidth of only 0.19%. This characteristic makes it more suitable for applications requiring high-Q narrowband antennas, precise measurement of frequency shifts such as wireless sensing.

Mason circuit model:
The Mason equivalent circuit model was developed to analyze the electrical properties of the PE μ-antenna. The circuit consists of layers arranged from top to bottom: the radiating layer, protective layer, top electrode, piezoelectric layer, bottom electrode, and seed layer. In this model, the characteristic acoustic impedance (Z) is defined as Z = E \cdot \rho, where E, , and β represent the Young's modulus, density, and acoustic propagation constant of the acoustic layer, respectively. The ZnO radiation layer is represented as an acoustic layer. Due to the piezoelectric effect, the acoustic impedance of the ZnO is modified accordingly to Z ra = (c E3 3 + e 23 3 / \epsilon S3 3) \cdot \rho. Additionally,  = \beta \cdot l , with l being the thickness of the acoustic layer. The piezoelectric layer is treated as a transformer circuit responsible for converting mechanical energy and electric one. Key parameters include k 2t = e 23 3 / (\epsilon S3 3 \cdot c E3 3), the static capacitance C 0 = \epsilon S3 3 \cdot A / d, the electromechanical coupling coefficient and the characteristic acoustic impedance of the piezoelectric layer Z pe = (c E3 3 + e 23 3 / \epsilon S3 3) \cdot \rho, where A is the FBAR effective region area and d is the thickness of the piezoelectric layer. Moreover, , S3 3  , e 3 3 and c E3 3 represent the piezoelectric layer's density, dielectric constant, piezoelectricity coefficient, and stiffness factor, respectively.

Finite element modeling of PE μ-antenna:
We developed a 3D finite element model with an active region area matching the device's structure in COMSOL (COMSOL Multiphysics 6.2). The simulations were conducted utilizing the Electrostatics and Solid Mechanics modules, with the incorporation of Piezoelectric Multiphysics field modules applied to the AlScN and ZnO layers. A voltage signal with an amplitude of 10 mV was applied to both the lower surface of the upper electrode and the upper surface of the lower electrode. To minimize the impact of transverse parasitic modes, the device was enclosed by a perfectly matched layer. In the simulations, a cell with an area of 11 μm2, as shown in Supplementary Fig. S6, was configured with layers the same as the antenna's structure. Periodic boundary conditions were implemented at the four boundaries (front, back, left, and right) of this cell. In the 2D simulation of the HBAR-PE μ-antenna, as depicted in Fig. 4a, the cavity beneath the AlN seed layer was filled with PSG to create the HBAR structure without the need for a cavity structure. Detailed material parameters for the layers employed in the simulations can be found in Supplementary Note 5.

Measurements:
(1) RF characterization: The S-parameters were measured using vector network analyzers (E5080B, Keysight, USA; ZNLE3, Rohde & Schwarz, Germany) according to measurement requirements, after standard short–open–load–through (SOLT) calibration. For antenna gain measurements, a horn antenna with known gain and the PE μ-antenna under test were connected to the calibrated ports of the VNA. The PE μ-antenna was contacted using a GSG probe on an RF probe station. The output power of VNA port was set to −10 dBm, and measurements were conducted at varying antenna separation distances. The impedance characteristics of the PE μ-antenna were converted from its reflection coefficient (S ). 22
(2) Wireless temperature sensing evaluation: The temperature response of the wireless sensing unit was evaluated using a wireless temperature sensing setup. The unit was fixed on a programmable heating plate and heated to target temperatures. The VNA was used to connect the unit and the receiving antenna for recording the S-parameters. The VNA port output power was set to –10 dBm, and the frequency resolution was 14 kHz.
(3) Wireless strain sensing evaluation: The strain response of the wireless sensing unit was characterized using a wireless strain sensing setup. The unit was bonded to a strain plate made of TC4 titanium alloy using strain gauge adhesive. To calibrate the applied strain, a resistive strain gauge was attached to the same plate. The unit was positioned adjacent to the strain gauge, aligned with its center, to ensure consistent strain exposure. The VNA was used to acquire the S-parameters by connecting the unit and the receiving antenna. The VNA port output power was set to –10 dBm, and the frequency resolution was 2.5 kHz.

Peer review rounds
[{"round": 1, "reviewer_comments": {"reviewer_1": "The authors present an acoustic-resonance-driven piezoelectric microantenna based on film and high-overtone bulk acoustic resonator (FBAR/HBAR) structures, with a total footprint of 0.019 mm². They demonstrate its wireless sensing capability with a transmission distance of up to one meter. This work addresses two fundamental challenges in wireless sensing: antenna miniaturization and the integration of sensing functionality within an ultracompact form factor. The research establishes a scalable and highly miniaturized device platform that ranks among the smallest wireless sensing units with an integrated antenna reported to date. As such, it is likely to attract significant interest in the fields of MEMS, wireless communication, and microscale sensing.\nIn my view, this represents a notable achievement in tackling long-standing challenges pursued by researchers and engineers. The technology holds promises for broad implications and applications across various domains, including industry, biomedicine (particularly implants), and space exploration. I would recommend consideration for publication in Nature Communications, pending the consideration for the following technical points.\n1. The authors need to ensure consistent terminology for the μ-antenna throughout the manuscript. The terms FBAR/HBAR-PE antenna and FBAR/HBAR-PE μ-antenna are used interchangeably (e.g., lines 89, 307, and 311). A single, consistent form would be better and clearer to readers.\n2. Fig. S2 and Fig. S3 show the characterization of the piezoelectric materials, but no comparison with previous studies is provided to indicate the material performance.\n3. In Fig. 3b, the antenna S22 exhibits two resonance peaks, but only one resonance produces an observable electromagnetic radiation peak in Fig. 3d. The reason for this discrepancy is not explained, and its impact on antenna performance should be discussed.\n4. The HBAR-PE μ-antenna exhibits a longer transmission distance than the FBAR-PE μ-antenna (Fig. 4), but the manuscript does not clarify how this improvement correlates with the increase in Q factor.\n5. The FBAR-PE μ-antenna could operate at both the first and second resonance modes. In Fig. 4, the HBAR-PE μ-antenna shows resonance enhancement at the first FBAR mode. Can it also operate at the second FBAR mode and generate the corresponding electromagnetic radiation for wireless sensing?\n6. Both the HBAR-ME and HBAR-PE μ-antennas are built on the same HBAR platform, but the manuscript does not provide a rationale for why their performance can be directly compared.\n7. In Fig. S12b, the comparison with Fig. 4d is presented, and the manuscript states that “the use of a PSG layer with lower acoustic impedance results in more effective reflection of acoustic stress back into the ZnO+FBAR region.” This statement does not explain the underlying mechanism. The authors should clarify the role of the PSG layer and how it influences acoustic stress reflection within the device.\n8. The authors demonstrate that the HBAR-based PE μ-antenna exhibits wireless sensing functionality, with the wireless transmission enabled by the antenna itself. However, it is not clear which parts of the structure are responsible for the sensing function.", "reviewer_2": "Miniaturized on-chip sensor units with integrated antennas remain a key technical challenge in modern microscale sensing, owing to the stringent constraints on device footprint and the demand for reliable wireless operation in complex environments such as wearable electronics, compact industrial systems, and distributed sensor networks. Critical design considerations include achieving antenna miniaturization without compromising sensing performance-ideally through vertical integration, establishing theoretical guidance for the design of nanoantennas capable of long-range wireless transmission. Despite significant progress in microscale sensing technologies, it remains difficult to simultaneously achieve compact device dimensions and high-performance wireless operation.\nThe manuscript reports a microantenna and a wireless microsensor with an active area of merely 0.02 mm2, effectively addressing many of the aforementioned challenges. The ultracompact sensor unit demonstrates both temperature and strain sensing capabilities, with a remarkable transmission distance of up to one meter among the best results reported to date. Furthermore, the integrated sensor unit exhibits excellent scalability, suggesting strong potential for diverse application scenarios. Compared with magnetoelectric (ME) film-based antennas, ZnO piezoelectric (PE) thin-film nanoantennas offer advantages in terms of material availability, ease of fabrication, and performance. Overall, this study lays an important foundation for the next generation of wireless micro-sensing systems.\nI would recommend publication in Nature Communications after the authors carefully address the following points:\n(1) HBARs exhibit multiple resonance peaks with high Q-factors. Can each resonance mode be utilized for sensing or wireless transmission? Do all peaks exhibit comparable performance? How can users select the most appropriate mode for specific applications? Moreover, is there a strategy to enhance the dominant resonance (e.g., increasing its Q-factor) while suppressing parasitic modes, thereby concentrating the acoustic energy for longer transmission distances? In short, could the HBAR-based nanoantenna design be further optimized for higher performance?\n(2) For large-scale fabrication, what are the observed variations in resonance frequency and Q-factor among fabricated HBARs? How do these variations influence the performance consistency of the nanoantennas and sensor units? The authors are encouraged to comment on the uniformity of ZnO-based antennas and HBARs both across a single wafer and between different wafers.\n(3) Although the demonstrated transmission distance reaches up to one meter, could the authors discuss possible strategies to extend the wireless communication range further? Insights into design, material, or system-level improvements that could enable longer-distance operation would be valuable.\n(4) While FBARs and HBARs have been demonstrated for temperature and strain sensing, they are, in principle, capable of detecting multiple physical parameters. What additional sensing functionalities could be realized with this microsensor platform? Furthermore, could the authors comment on the feasibility of adapting the proposed device for biomedical or implantable applications, which would significantly broaden its impact?\n(5) Figures S2 and S3 present the characterization of the piezoelectric materials. The manuscript would be strengthened if these results were compared with those from previous studies. Any observed improvement in ZnO film quality could directly enhance the overall performance of the sensor/antenna unit.\n(6) PSG is used as an acoustic reflector, though it increases fabrication complexity. Is the inclusion of the PSG layer essential? How would the device performance change if a silicon substrate were used directly as the acoustic reflector?\n(7) Although ZnO offers relatively good piezoelectric performance, its piezoelectric coefficient (d33) is lower than that of some alternative materials. Would it be feasible to employ higher-performance materials such as PZT with larger d33 and electromechanical coupling coefficients? What are the key challenges or limitations that prevent the integration of such materials into the nanoantenna design?"}, "author_response": "Response Letter\nMulti-mode piezoelectric radiation-based microantennas and miniaturized wireless sensing\nunit driven by bulk acoustic waves (NCOMMS-25-79248A)\nWe would like to express our sincere gratitude to all the reviewers for their positive comments on\nour manuscript, and their valuable and constructive suggestions for the revision. In the following\nsection, we address each of the issues raised by the reviewers and outline the corresponding\nchanges we have made in our revised manuscript. We believe that implementing the reviewers’\nsuggestions has significantly enhanced the quality of our work. Our aim is to ensure that the revised\nmanuscript meets the criteria of impact, innovation, and interest for publication in Nature\nCommunication, as judged by both the reviewers and the editor.\nReviewer #1:\nThe authors present an acoustic-resonance-driven piezoelectric microantenna based on film and\nhigh-overtone bulk acoustic resonator (FBAR/HBAR) structures, with a total footprint of 0.019\nmm². They demonstrate its wireless sensing capability with a transmission distance of up to one\nmeter. This work addresses two fundamental challenges in wireless sensing: antenna\nminiaturization and the integration of sensing functionality within an ultracompact form factor.\nThe research establishes a scalable and highly miniaturized device platform that ranks among the\nsmallest wireless sensing units with an integrated antenna reported to date. As such, it is likely to\nattract significant interest in the fields of MEMS, wireless communication, and microscale sensing.\nIn my view, this represents a notable achievement in tackling long-standing challenges pursued by\nresearchers and engineers. The technology holds promises for broad implications and applications\nacross various domains, including industry, biomedicine (particularly implants), and space\nexploration. I would recommend consideration for publication in Nature Communications,\npending the consideration for the following technical points.\nand encouraging comments on our work. We have carefully addressed all the technical points\nraised, particularly those concerning device design and experimental validation, and revised the\nmanuscript accordingly. The changes are highlighted in blue. We believe these revisions have\nimproved the clarity and quality of the paper.\nComment 1. The authors need to ensure consistent terminology for the μ-antenna throughout the\nmanuscript. The terms FBAR/HBAR-PE antenna and FBAR/HBAR-PE μ-antenna are used\ninterchangeably (e.g., lines 89, 307, and 311). A single, consistent form would be better and clearer\nto readers.\nResponse: We are grateful for the reviewer for pointing out the inconsistency in terminology. We\nhave standardized the terminology across the manuscript to enhance clarity for readers. All\ncorresponding changes are marked in blue in the revised manuscript.\nComment 2. Fig. S2 and Fig. S3 show the characterization of the piezoelectric materials, but no\ncomparison with previous studies is provided to indicate the material performance.\nResponse: We appreciate the reviewer very much for this helpful suggestion. Figs. S2 and S3\nverify the crystallographic quality and orientation consistency of the deposited piezoelectric films,\nensuring that their properties satisfy the requirements for the device design. The AlScN films\nsimilarly demonstrate high crystallinity, with (0002) FWHM superior to previously reported\nAlScN-based studies (Supplementary References [1–5]). The ZnO films, prepared without any\npost-deposition annealing, exhibit a preferential c-axis orientation, with an XRD (002) FWHM of\n0.61° and grain sizes of 150–200 nm, while the piezoelectric coefficient approaches standard ZnO\nvalues. These metrics demonstrate higher crystallographic quality compared with previously\nreported values (FWHM: 0.19–2.45°, most grain sizes <100 nm, Supplementary References [6–\n10]).\nThese results confirm that the deposition processes reliably yield high-quality piezoelectric layers,\nproviding a robust foundation for the HBAR-PE μ-antenna to achieve both high performance and\nprocess reproducibility. The descriptions of Figs. S2 and S3 have been updated in the\nSupplementary Information, and relevant comparative references have been added.\nproduces an observable electromagnetic radiation peak in Fig. 3d. The reason for this discrepancy\nis not explained, and its impact on antenna performance should be discussed.\nResponse: We sincerely thank the reviewer for raising this insightful question. The first resonance\npeak observed in the measured S spectrum (Fig. 3b) corresponds to a parasitic acoustic mode\n22\ninduced by fabrication-related imperfections, such as minor structural asymmetries or film\nnonuniformities. This parasitic resonance does not appear in simulation because the model\nassumes ideal geometries and homogeneous material properties.\nImportantly, this parasitic mode does not contribute to electromagnetic radiation. The radiation\ncoupling of the μ-antenna is predominantly governed by the thickness resonance mode, which\ndefines the primary acoustic–electrical conversion pathway. Both the simulation (Fig. 2b) and the\nmeasured radiation spectrum (Fig. 3d) confirm the absence of any radiation peak corresponding\nto this parasitic mode. Furthermore, since its resonance frequency is well separated from that of\nthe dominant mode, the parasitic response has a negligible impact on the overall antenna\nperformance. We have added a brief discussion in the manuscript.\nComment 4. The HBAR-PE μ-antenna exhibits a longer transmission distance than the FBAR-PE\nμ-antenna (Fig. 4), but the manuscript does not clarify how this improvement correlates with the\nincrease in Q factor.\nResponse: We sincerely appreciate the reviewer’s valuable comment. For the HBAR-PE μ-\nantenna, the HBAR configuration provides a high-Q platform that fundamentally enhances the\nantenna performance. The higher Q factor reduces energy dissipation around the resonant\nfrequency, enabling more efficient acoustic–electrical energy conversion and greater radiation\nintensity. As a result, the HBAR-PE μ-antenna exhibits a gain improvement of approximately 4-\n6.7 dBi compared with the first-order mode of the FBAR-PE antenna.\nMoreover, the high-overtone thickness modes supported by the HBAR structure possess an\nintrinsically small f , which effectively confines the acoustic energy within individual resonant\nr\nmodes. This leads to a narrower resonance bandwidth (0.19%) and stronger spectral concentration,\nthereby improving the signal-to-noise ratio at the resonance frequency. Consequently, the extended\ntransmission distance of the HBAR-PE μ-antenna can be attributed to these combined effects—its\n3\nenhanced radiation efficiency and improved signal quality arising from the higher Q factor. We\nhave added a brief discussion in the discussion of Fig. 4i to clarify this correlation.\nComment 5. The FBAR-PE μ-antenna could operate at both the first and second resonance modes.\nIn Fig. 4, the HBAR-PE μ-antenna shows resonance enhancement at the first FBAR mode. Can it\nalso operate at the second FBAR mode and generate the corresponding electromagnetic radiation\nfor wireless sensing?\nZnO AlScN 1\nSi PSG Si\ny Cross-sectionSi\nx Line\n1\n50\n60\n70\nFig. R1. HBAR-PE μ-antenna in the second operating band. a Normalized stress distribution across each layer of\nthe antenna at a resonant frequency within the operating band, showing the resonance enhancement effect. b\nNormalized stress distribution along the cross-section line indicated in a for a resonant mode within the operating\nband and a mode outside the operating band, where the out-of-band stress level exceeds that in the first operating\nband, indicating weakened in-band resonance confinement. c and d S of the AlScN-based HBAR-PE μ-antennas\n22\nwith 9.6% and 20% Sc contents, respectively. e and f Transmission coefficients (S and S ) of the AlScN-based\n21 12\nHBAR-PE μ-antennas with 9.6% and 20% Sc contents, respectively.\nResponse: We highly appreciate the reviewer’s insightful comment and apologize for not\nproviding a clear explanation in the original manuscript. To clarify, we have now included\nadditional simulations and experimental measurements for the HBAR-PE μ-antenna in the second\noperating band (Fig. R1).\nAs shown in Fig. R1a and b, within the second operating band, resonance enhancement still occurs,\nwith localized stress increase in the FBAR region under in-band resonant modes. Experimental\nresults in Fig. R1c-f confirm the existence of this second operating band, corresponding to the\nsecond FBAR counterpart resonance. However, at higher frequencies, the acoustic wavelength\nshortens, leading to increased standing waves in the substrate and enhanced out-of-band energy\n4\n)Bd(21S12S\n50 S21\nS12 Ga  i  nm m a a x x    0 3 .0 7 1 .1 3 8 % dBi\n60\n70\n3.8 4.0 4.2\nFreq. (GHz)\n)Bd(21S12S\nS21\nGainmax33.76 dBi S12  max0.028%\n0 2 4 6 8 3.8 4.0 4.2\nDistance from Top Surface (m) Freq. (GHz)\nssertS\ndesilamroN\nQ Km e a ff x2   6 0 3 .3 2 6%\nOut-of-band 0.5\n0\n0.5\nZnO PSG Si\n1 FBAR\n)Bd(22S\n0\n3\n6\n9\n12\n)Bd(22S\na c d\n0\n3\n6\n9 Q Km e a ff x2   4 0 7 .5 2 4%\n12\nb e f\nleakagewith stress levels at out-of-band resonance points exceeding those out of the first\noperating band. Combined with increased boundary scattering and energy leakage, these factors\ncollectively cause higher losses and a significant reduction in the in-band Q factor. Ultimately,\nalthough resonance enhancement is still observed within the second operating band, the antenna’s\nradiation efficiency and gain are substantially reduced, rendering it unsuitable for effective\nwireless signal transmission.\nTherefore, for practical wireless sensing, only the first resonance mode is used, ensuring effective\nradiation and stable signal transmission. We have updated the manuscript to describe this mode\naccordingly, and Fig. R1 has been added to the Supplementary Information as Fig. S14.\nComment 6. Both the HBAR-ME and HBAR-PE μ-antennas are built on the same HBAR\nplatform, but the manuscript does not provide a rationale for why their performance can be directly\ncompared.\nResponse: We sincerely appreciate the reviewer’s thoughtful comment. Both the HBAR-ME and\nHBAR-PE μ-antennas were fabricated on the same AlScN-based HBAR platform using the same\nprocess except the processes for PE and ME antennas separatedly, as shown in Fig. S4. The key\nfabrication steps—including AlScN thin-film deposition, patterning, and electrode formation—are\nconsistent across both devices. Unlike air-cavity FBAR structures, the solid-substrate HBAR\nconfiguration ensures superior mechanical stability and effectively minimizes acoustic losses\narising from stress relaxation or film deformation.\nIn this process, the FeGaB magnetostrictive layer of the HBAR-ME μ-antenna was directly\ndeposited on the AlScN piezoelectric layer, maintaining an identical device geometry and acoustic\npropagation path to that of the HBAR-PE μ-antenna. As shown in Fig. R2, the fabricated AlScN-\nbased HBAR devices which are the basic structures for HBAR-ME and HBAR-PE μ-antennas,\nconsistently exhibit high-Q resonances with comparable Q factors and f in both antenna types.\na\nTherefore, the HBAR-ME and HBAR-PE μ-antennas share equivalent fabrication processes,\nstructural configurations, and acoustic characteristics. The observed difference in radiation\nbehavior originates from their distinct radiation mechanisms—piezoelectric versus\nmagnetostrictive—thereby rendering the performance comparison well justified. We have updated\nthe manuscript accordingly, and Fig. R2 has been added to the Supplementary Information as Fig.\nS15.\n5\n0\n5\n10\n15\n20\n3 4 5 6 7 4.95 5.05 5.15\nFreq. (GHz) Freq. (GHz)\nFig. R2. Measured reflection coefficient (S ) spectrum of the AlScN-based HBAR device.\n“the use of a PSG layer with lower acoustic impedance results in more effective reflection of\nacoustic stress back into the ZnO+FBAR region.” This statement does not explain the underlying\nmechanism. The authors should clarify the role of the PSG layer and how it influences acoustic\nstress reflection within the device.\nResponse: We would thank the reviewer for the comments very much. As shown in Fig. S12b,\nunder the in-band resonance mode, when the PSG layer is absent, the acoustic waves generated\nwithin the FBAR region directly propagate into the Si substrate, reducing the acoustic stress within\nthe ZnO+FBAR region to approximately 50% of that observed in the PSG-based design. This\nindicates a weakened in-band resonance effect.\nThe underlying mechanism is associated with acoustic impedance mismatch. The PSG layer\npossesses a lower acoustic impedance (Z =1.25107 kg/(m2s)) compared with the Si substrate\nPSG\n(Z =1.96×107 kg/(m2s)). Therefore, at the AlN seed layer/PSG interface, a high-to-low acoustic\nSi\nimpedance boundary is formed, which reflects a greater portion of the downward-propagating\nacoustic waves back into the ZnO+FBAR region. This enhanced reflection effectively confines\nthe acoustic energy within the active region and strengthens the in-band resonance response.\nFurthermore, the description of Fig. S12 in the Supplementary Information has been updated\naccordingly to clarify this mechanism in greater detail.\nComment 8. The authors demonstrate that the HBAR-based PE μ-antenna exhibits wireless\nsensing functionality, with the wireless transmission enabled by the antenna itself. However, it is\nnot clear which parts of the structure are responsible for the sensing function.\nResponse: We sincerely thank the reviewer for pointing out the missing explanation and feel sorry\nthat we did not explain the sensing mechanism clearly. In the HBAR structure, the sensing\n6\n)Bd(11S\n0\n5\n10\n15\n20\n)Bd(11S\nfa= 17.2 MHz\nQmax= 1251\nfunctionality is primarily determined by the multilayer stack. The response of a bulk acoustic\nresonator to physical quantities such as temperature and strain originates from variations in the\nmechanical and acoustic properties of the constituent materials (e.g., elastic modulus, density, and\nacoustic velocity) under environmental perturbations. These variations modify the propagation\nvelocity and path of acoustic waves within the resonant cavity, resulting in a measurable resonance\nfrequency shift.\nSpecifically, in the HBAR structure, temperature and strain affect the elastic constants and\ndensities of the multilayer stack. The resonance frequency can be calculated according to Eq. (1)\nin the manuscript, with the PSG layer and Si substrate included in the layer structure. Since the Si\nsubstrate accounts for the majority of the total thickness in the HBAR stack, the longitudinal\nacoustic velocity of the HBAR can be approximated as that of the Si substrate. This is because the\nmultilayer structure of the HBAR can be regarded as a series of springs, and its effective elastic\nconstant can be calculated as follows:\n(1)\n1 1 ∑𝑗𝑑𝑗\nFor the HBAR, the subs 𝐸 tr 𝑒 a 𝑞 te = th 𝑠 i 𝑒 c 𝑓𝑓 kn = es ∑ s 𝑖 ∑i𝑗s 𝑑 𝑑 𝑖 m𝑗𝑐3u 1 3c,𝑖h = g ∑ re 𝑖𝑐a3 𝑑 t3 𝑖 e,𝑖r than that of the other functional layers;\ntherefore, the effective elastic modulus of the device is mainly determined by the elastic modulus\nof the substrate. In other words, the mechanical and acoustic properties of the silicon substrate play\na main role in determining the resonance frequency characteristics of the device. This conclusion\nis also supported by the experimental results: the f of the HBAR-ME μ-antenna, the HBAR-PE\na\nμ-antenna, and the HBAR device shown in Fig. R2 are all approximately 17 MHz, as they have\nthe same silicon substrate. Hence, the strain and temperature sensitivities of the silicon substrate\ndirectly determine the overall sensing characteristics of the HBAR-based wireless sensing unit.\nWe have added this discussion as Supplementary InformationNote 2.\nReviewer #2:\nMiniaturized on-chip sensor units with integrated antennas remain a key technical challenge in\nmodern microscale sensing, owing to the stringent constraints on device footprint and the demand\nfor reliable wireless operation in complex environments such as wearable electronics, compact\n7\nindustrial systems, and distributed sensor networks. Critical design considerations include\nachieving antenna miniaturization without compromising sensing performance-ideally through\nvertical integration, establishing theoretical guidance for the design of nanoantennas capable of\nlong-range wireless transmission. Despite significant progress in microscale sensing technologies,\nit remains difficult to simultaneously achieve compact device dimensions and high-performance\nwireless operation.\nThe manuscript reports a microantenna and a wireless microsensor with an active area of merely\n0.02 mm2, effectively addressing many of the aforementioned challenges. The ultracompact sensor\nunit demonstrates both temperature and strain sensing capabilities, with a remarkable transmission\ndistance of up to one meter among the best results reported to date. Furthermore, the integrated\nsensor unit exhibits excellent scalability, suggesting strong potential for diverse application\nscenarios. Compared with magnetoelectric (ME) film-based antennas, ZnO piezoelectric (PE)\nthin-film nanoantennas offer advantages in terms of material availability, ease of fabrication, and\nperformance. Overall, this study lays an important foundation for the next generation of wireless\nmicro-sensing systems.\nI would recommend publication in Nature Communications after the authors carefully address the\nfollowing points:\nResponse to the reviewer’s overall assessment: We sincerely appreciate the reviewer’s valuable\nsuggestions and positive recognition of our work. We have carefully addressed all the points raised\nand revised the manuscript accordingly. Detailed responses to each comment are provided below.\nComment 1. HBARs exhibit multiple resonance peaks with high Q-factors. Can each resonance\nmode be utilized for sensing or wireless transmission? Do all peaks exhibit comparable\nperformance? How can users select the most appropriate mode for specific applications? Moreover,\nis there a strategy to enhance the dominant resonance (e.g., increasing its Q-factor) while\nsuppressing parasitic modes, thereby concentrating the acoustic energy for longer transmission\ndistances? In short, could the HBAR-based nanoantenna design be further optimized for higher\nperformance?\nResponse: We sincerely thank the reviewer for this thoughtful and constructive comment. The\nHBAR structure indeed supports multiple high-Q resonance modes; however, only the modes\nlocated within the operational bandwidth of the HBAR (i.e., within the envelope peak of S ) can\n22\n8\neffectively enable wireless transmission. This is because the in-band modes experience a\n“resonance enhancement” effect, where the stress amplitude is significantly amplified, thereby\nimproving the acoustic–electrical radiation efficiency. For mode selection, we recommend using\nthe radiation strength as the primary criterion. Typically, the resonance modes located near the\ncenter of the in-band region exhibit superior impedance matching and higher radiation gain. The\namplitudes of other resonance modes become smaller when they are away from the central mode\nand eventually become diminished. Therefore, it is clear that it is better to select the resonance\nmode located near the center of the in-band region for sensing with longer transmission range.\nRegarding performance optimization, suppressing out-of-band modes is an effective strategy. In\nthis work, we achieved this by introducing a PSG layer into the HBAR structure. As shown in Fig.\n4d and Fig. S12b, the PSG layer effectively reduces the stress amplitude of the out-of-band modes,\nthereby minimizing the leakage of acoustic energy from the in-band region and concentrating the\nenergy to enhance in-band radiation. This approach is conceptually analogous to the use of Bragg\nreflector layers in FBAR designs, where alternating high/low acoustic impedance layers confine\nthe acoustic energy within the desired resonance band. As an extreme example, if the effective\nacoustic impedance of the PSG region were further reduced to nearly zero—such as by forming\nan air cavity—the out-of-band resonances could be completely suppressed, causing the HBAR\nstructure to effectively transition into an FBAR configuration.\nAdditionally, other optimization routes—such as employing low-acoustic-loss substrates, high-\nquality piezoelectric films, or impedance-matching circuitry—are also viable and will be explored\nin future work. We have added some discussions in the Supplementary Information Note 3.\nComment 2. For large-scale fabrication, what are the observed variations in resonance frequency\nand Q-factor among fabricated HBARs? How do these variations influence the performance\nconsistency of the nanoantennas and sensor units? The authors are encouraged to comment on the\nuniformity of ZnO-based antennas and HBARs both across a single wafer and between different\nwafers.\nResponse: We are grateful for the reviewer for raising this important point. The HBAR structures\nin this work were fabricated using a mature MEMS process, and the measured resonant\ncharacteristics are in agreement with finite-element simulations, confirming the reliability of both\nthe structural design and fabrication protocol. The critical layers of the HBARs, including the\n9\npiezoelectric and electrode films, have been extensively validated on production-ready process\nplatforms, ensuring high process stability and reproducibility.\nZnO piezoelectric layers were subsequently deposited on the HBAR stack by magnetron sputtering\nat the university’s microfabrication Center. The films also exhibited a stable c-axis orientation and\nuniform crystalline quality across various substrates, including Si shown in Fig. S3 and Al, Mo,\nand SiO shown in Fig. R3. XRD measurements show that the FWHM variation of the (002)\n2\ndiffraction peak remains typically within 5%, indicating uniformity of the ZnO layer across wafers\nand deposition batches.\nDevice-to-device variations in resonance frequency and Q-factor are primarily attributed to minor\ndeviations in film thickness and interfacial roughness. Given the maturity and repeatability of the\nMEMS and ZnO deposition processes, these variations are small and have negligible influence on\nsystem-level performance metrics such as radiation efficiency and sensing response. Overall, both\nintra-wafer and inter-wafer comparisons confirmed that the ZnO-based HBAR-PE μ-antennas\nexhibit consistent resonance and Q characteristics. These results demonstrated the high\nreproducibility and scalability of the fabrication process, supporting its applicability to large-scale\nmanufacturing and integrated sensing systems.\nMo (110)\n2() 1\nFig. R3. XRD patterns of 500 nm ZnO films deposited on Al, Mo, and SiO films atop Si substrates. The (002)\n2\ndiffraction peaks indicate a stable c-axis orientation and uniform crystalline quality across all samples.\nComment 3. Although the demonstrated transmission distance reaches up to one meter, could the\nauthors discuss possible strategies to extend the wireless communication range further? Insights\ninto design, material, or system-level improvements that could enable longer-distance operation\nwould be valuable.\nResponse: We would thank the reviewer very much. This is a good point raised by this reviewer.\nThe current results demonstrate that the HBAR-PE μ-antenna enables stable wireless sensing\nsignal transmission over a distance of approximately one meter, confirming its efficient acoustic–\n10\n).u.a(\nytisnetnI\nZnOon SiO2film\nZnOon Al film\nZnOon Mo film\nZnO(002)\n30 35 40 45\nelectric conversion performance. To further extend the wireless transmission range, several\nstrategies can be considered:\n1. Material optimization: Improving the thin-film deposition process to reduce acoustic losses\nand employing piezoelectric films with higher electromechanical coupling coefficients in the\ndual piezoelectric radiation layers can enhance energy conversion efficiency. In addition,\nselecting low-loss substrate materials can minimize acoustic energy dissipation within the\nHBAR structure.\n2. Structural design improvements: Increasing the effective radiating area, optimizing the HBAR\ncavity thickness and acoustic reflection interfaces, and introducing high-Q design concepts\n(such as Mo ring structures used in FBARs) could further improve the resonance quality factor\nand radiation efficiency.\n3. System-level enhancement: Employing impedance-matching networks or antenna array\nconfigurations can boost radiated power, and improving the receiver-side performance through\nhigher sensitivity, low-noise amplification, and advanced signal processing to enhance the\noverall signal-to-noise ratio.\nThese strategies are expected to further extend the wireless communication range, and we are\nactively exploring and validating them in ongoing work.\nWe have added this discussion in the Supplementary Information as Note 3.\nComment 4. While FBARs and HBARs have been demonstrated for temperature and strain\nsensing, they are, in principle, capable of detecting multiple physical parameters. What additional\nsensing functionalities could be realized with this microsensor platform? Furthermore, could the\nauthors comment on the feasibility of adapting the proposed device for biomedical or implantable\napplications, which would significantly broaden its impact?\nResponse: We genuinely thank the reviewer for this valuable and forward-looking comment.\nIndeed, both FBAR and HBAR platforms exhibit intrinsic sensitivity to perturbations in their\nsurrounding environment, making them promising candidates for a broad range of physical,\nchemical, and biological sensing applications as demonstrated by many FBAR-based physical and\nbiochemical sensing. The extensibility of the sensing functionalities of present HBAR-PE μ-\nantenna sensor can be categorized as follows:\n1. Mass sensing: Based on the mass-loading effect, adsorption of gas molecules, liquid\n11\ncomponents, or biomolecules on the resonator surface induces measurable resonance\nfrequency shifts, enabling highly sensitive detection of molecular adsorption, binding, and\nreaction processes.\n2. Temperature and strain sensing: Variations in temperature or mechanical strain modify the\nelastic constants and internal stress distribution within the piezoelectric film and the supporting\nlayers, resulting in distinct resonance frequency shifts that allow precise monitoring of thermal\nand mechanical conditions.\n3. Electric and magnetic field sensing: For piezoelectric films exhibiting ferroelectric properties,\nexternal electric fields or variations in the dielectric environment can modulate polarization\nand material permittivity, thereby altering the resonant characteristics. In addition, by\nintegrating magneto-responsive thin films, magnetic field detection can be achieved through\nmagneto-elastic or magneto-electric coupling effects.\n4. Chemical and biosensing: By functionalizing the resonator surface with selective adsorption\nlayers or biorecognition molecules, the device can respond specifically to targeted chemical\nspecies or biomolecular markers, allowing label-free and highly selective detection.\nFurthermore, the proposed HBAR-PE μ-antenna features dimensions smaller than most reported\nimplantable sensors and inherently combines wireless transmission with sensing capability.\nPrevious studies (e.g., Ref. [36]) have demonstrated that ME antennas can enable wireless energy\ntransfer for implantable systems. Building on this foundation, the PE μ-antenna developed in this\nwork offers higher radiation efficiency, suggesting feasibility for wireless implantable sensing in\nbiological environments. For future biomedical applications, biocompatible encapsulation could\nbe employed to enhance long-term operational stability under physiological conditions, while\nfurther improving the Q-factor (e.g., >5000) would enhance signal readability and enable fully\nwireless, passive implantable sensing. We are extensively working on some of these.\nWe have added a discussion in the Supplementary Information as Note 4.\nComment 5. Figures S2 and S3 present the characterization of the piezoelectric materials. The\nmanuscript would be strengthened if these results were compared with those from previous studies.\nAny observed improvement in ZnO film quality could directly enhance the overall performance\nof the sensor/antenna unit.\nbeen added in the Supplementary Information, demonstrating that the AlScN, ZnO and FeGaB\nfilms prepared here achieve state-of-the-art quality. Reduced loss in the ZnO films (Fig. S7b) is a\nkey factor enhancing antenna radiation performance. Key ZnO characterization metrics directly\nrelated to antenna performance include:\n1. XRD crystal orientation: highly preferential c-axis orientation ensures efficient piezoelectric\nenergy conversion.\n2. FWHM: narrower peaks indicate higher crystallinity and fewer defects.\n3. Grain size: larger grains reduce grain boundaries and internal defects, enhancing mechanical\nstability and piezoelectric efficiency.\n4. Piezoelectric coefficient (d ): higher values directly translate to stronger electromechanical\n33\ncoupling and improved radiation efficiency.\n5. Surface roughness: smoother surfaces improve electrode contact uniformity and reduce\nacoustic scattering losses.\nThese metrics collectively demonstrate that the high crystallinity and superior piezoelectric\nproperties of the ZnO films contribute directly to the enhanced performance of the HBAR-PE\nantenna and sensor units, underlining the material advantage in this work.\nComment 6. PSG is used as an acoustic reflector, though it increases fabrication complexity. Is\nthe inclusion of the PSG layer essential? How would the device performance change if a silicon\nsubstrate were used directly as the acoustic reflector?\n1\n0.5\n0\n0.5\n0 2 4 6 8\nDistance from Top Surface (m)\nFig. R4. Normalized stress distribution of the HBAR-PE μ-antenna.a Resonant mode within the operational band.\n13\nssertSdesilamroN\nWith PSG layer\nWithout PSG layer\n0.02\n0\n0.02\nZnO+\nFBAR Substrate\n0 2 4 6 8\nDistance from Top Surface (m)\nssertSdesilamroN\na b\nWith PSG layer\nWithout PSG layer\nZnO+\nFBAR Substrate\nResponse: We are grateful for this valuable question. The inclusion of the PSG layer indeed\nincreases the fabrication complexity; however, it was introduced for two primary reasons:\n1. Process compatibility: The PSG layer enhances the fabrication universality between FBAR\nand HBAR structures, allowing both devices to be realized through a unified MEMS process\nflow.\n2. Device performance: As shown in Fig. R4 (reorganized from Fig. 4d and Fig. S12b), the\nintroduction of the PSG layer does not alter the fundamental operating principle of the\nHBAR—namely, the in-band resonance enhancement that amplifies the stress amplitude\nwithin the FBAR region. However, it effectively confines the acoustic energy within the in-\nband resonant modes while suppressing out-of-band parasitic responses, thereby minimizing\nacoustic leakage into the substrate. According to Eqs. (2) and (3) in the manuscript,\n2\nand . Consequently, the improved acoustic confinement provided by the PSG layer\n𝑃𝑜𝑢𝑡 ∝ 𝑝0\nresults in approximately a fourfold increase in radiated power compared with a device using a\n𝑝0 ∝ 𝑻\nsilicon substrate as the acoustic reflector.\nWe have added some the comparison and explanations in Figs. S12 for clarification.\nComment 7. Although ZnO offers relatively good piezoelectric performance, its piezoelectric\ncoefficient (d33) is lower than that of some alternative materials. Would it be feasible to employ\nhigher-performance materials such as PZT with larger d33 and electromechanical coupling\ncoefficients? What are the key challenges or limitations that prevent the integration of such\nmaterials into the nanoantenna design?\nResponse: We would sincerely thank the reviewer for this important comment. While ZnO serves\nas a practical piezoelectric material for the present platform, the exploration of alternative high-\nperformance piezoelectric films provides a potential route to further enhance device capabilities,\nand the diversity of piezoelectric materials underpins the potential scalability of this research.\nPZT thin films exhibit the highest piezoelectric coefficients and electromechanical coupling\ncoefficients currently available, and in principle could replace ZnO to improve antenna radiation\nefficiency and signal strength. However, the fabrication of high-quality PZT thin films remains\nchallenging. Achieving low defect density and high crystallinity is difficult, and deviations from\nbulk material properties can significantly degrade the device quality factor (Q). Indeed, many\nreported PZT-based FBAR devices exhibit Q values below 200. Furthermore, commonly used PZT\n14\ndeposition methods—including high-temperature sintering, sol–gel processing, and sputtering on\nspecialized matching layers—are generally incompatible with established MEMS fabrication\nprocesses. Another intrinsic limitation is associated with the ferroelectric nature of PZT:\npolarization switching induces hysteretic losses, which further increases energy dissipation.\nMitigating these losses, analogous to improving the soft magnetic performance of magnetostrictive\nmaterials, remains an open challenge for integrating high-performance ferroelectric piezoelectrics\ninto MEMS devices.\nBy contrast, ZnO can be reliably deposited using mature, MEMS-compatible processes, yielding\nfilms with high crystallinity, low defect density, and reproducible piezoelectric properties as\ndemonstrated by our work. Consequently, ZnO represents a feasible and viable choice for the\ncurrent device platform. With continued progress in thin-film fabrication and integration\ntechniques, PZT or other high-performance piezoelectric materials may become viable alternatives,\noffering opportunities to further enhance the performance of HBAR-based antennas and sensor\nunits.\n15"}, {"round": 2, "reviewer_comments": {"reviewer_1": "The authors have substantially improved the manuscript in response to the previous review comments. All major concerns have been adequately addressed, and the revisions have significantly enhanced the overall quality of the paper. The paper uniquely presents an innovative class of piezoelectric μ-antennas that harness acoustic-wave–induced polarization rotation to enable efficient microscale radiation. The demonstrated PE μ-antennas achieve dual-frequency operation, while the HBAR-PE μ-antenna exhibits ultrahigh Q factors and improved radiation efficiency. Such developed integrated sensor platform is compact, low-power, and well aligned with emerging applications in implantable medical devices, wearable electronics, and aerospace sensing and communication systems. The contributions are technically compelling and of high relevance to researchers across multiple disciplines. In its revised form, the manuscript meets the standards for publication, and therefore, it is recommended for acceptance.", "reviewer_2": "Reviewer Comment on Revised Manuscript\nI have carefully reviewed the authors’ responses to my previous comments and examined the corresponding revisions. The authors have addressed all points thoroughly and constructively, providing additional simulations, experimental data, and clarifications that substantially strengthen the manuscript.\nTheir revisions not only resolve the technical queries but also enhance the depth, rigor, and readability of the work. The new figures and supplementary notes further support the conclusions and demonstrate the robustness of the proposed miniaturized wireless sensing platform.\nOverall, the revised manuscript presents a clearer, more complete, and compelling case for the novelty and impact of this research. I am satisfied with the quality of the revisions and believe the manuscript now meets the high standards of Nature Communications.\nI therefore recommend acceptance of the manuscript in its current form."}, "author_response": ""}]

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