Electromagnetic Simulation Software

Using EM Simulation for 5G Device Design and Network Planning
A Collection of Application Examples Featuring Remcom’s EM Simulation Software
Introduction
5G is pushing the boundaries of wirelesscommunications and wireless device
design. New innovations are neededin order to accommodate increased
requirements for Enhanced Mobile
Broadband, Massive Machine-Type
Communications, and Ultra-Reliable, Low
Latency Communications.
Remcom’s simulation products provide a complete solution, from system and antenna
design, through performance assessment in realistic, simulated environments, andplanning for deployment in 5G networks. Our mission is to provide accurate solutions,
optimized for performance to help our customers reliably predict how their systems will
behave in the real world. Through active research and development, we have beendeveloping new products and adding capabilities to stay at the forefront of modeling andsimulation technology for 5G and other advances in wireless communications.
We hope you enjoy this collection of application examples written by Remcom’s
engineers. They demonstrate how Remcom’s software can be used to solve challenges
related to 5G and MIMO use cases such as MIMO and array design, 5G urban small
cells, fixed wireless access, indoor WiFi and mmWave, beamforming, and more.

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Using EM Simulation for 5G Device Design and Network Planning

Table of Contents
Example 1: 5G Massive MIMO Outdoor Communications Analysis

Example 2: mmWave Channel Modeling with Diffuse Scattering inan Office Environment

Example 3: Simulation of MIMO Handset Antenna Array
Performance with Varying Hand Positions

Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna
Array for 5G

Example 5: Throughput of a 5G New Radio FD-MIMO System in an
Urban Area Using Custom Beamforming

Example 6: Combined 5G/4G Antenna Analysis in a Smartphoneusing EM Simulation

Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna
Array for 5G at 28 GHz

Example 8: Design and Simulation of 28 GHz Beamforming
System and Antenna Array for 5G Network Base Stations

Example 9: 5G Antenna Array Simulation for mmWave
Wireless Virtual Reality Headsets

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Example 1: 5G Massive MIMO Outdoor Communications Analysis
Summary: In this example the signal transmission between a massive MIMO basestation and a mobile device located in downtown Rosslyn is analyzed using Wireless
InSite’s MIMO capability.
To demonstrate the concept of MIMO simulation assessment, Wireless InSite’s MIMO
capability has been used to set up a small cell scenario in a dense urban environmentin Rosslyn, Virginia (Figure 1). The base station is positioned atop a post on a medianat a major intersection at a height of 10 meters above street level. For the purpose ofcomparison, the base station has been defined with two alternative configurations: (1)a single dipole antenna, and (2) a large MIMO antenna. Both use a carrier frequency of
28 GHz.
The imported city model and the terrain contain a total of approximately 3,800 faces.
The buildings and terrain are assigned concrete material properties with conductivityof 0.484 S/m and permittivity of 5.31. Figure 2 shows a close-up of the massive MIMO
antenna, defined as an 8x8 cross-polarized dipole array. The gain pattern of the firstelement is displayed in the array view. A mobile device is defined to move along a route
through the city and is shown approaching the base station location from the west, andthen turning to move away from it to the north.

Figure 1: Rosslyn 3D building and terrain data are set up
for SISO and MIMO analysis within Wireless InSte.

Figure 2: The 128-element Massive MIMO array is located ona Tx point several feet above the ground with a moving Rx set
in the downtown area.

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Example 1: 5G Massive MIMO Outdoor Communications Analysis

Wireless InSite MIMO contains an array builder which allows users to design an n-element sized MIMO array. This tool also gives users control over individual elements of
the array such as positioning, antenna pattern, and rotation. Figure 3 shows the 128-element array within the array builder. For simplicity, this array has been built with dipole
antennas, placed at half-wavelength spacing, and arranged with both vertically-polarized
and horizontally-polarized elements placed at each of the 64 positions shown. However,it should be noted that the software can handle many different types of analytical patterns and can also utilize imported patterns generated by high-fidelity array simulations
from a full-wave solver, such as Remcom’s XFdtd.

Figure 3: The 128-element Massive MIMO array was built in Wireless InSite’s powerful array builder.

Wireless InSite’s study area properties allows users to
configure and control various aspects of the physics
and ray-tracing calculations. Figure 4 shows the study
area setup dialogue for the massive MIMO simulation. The study area is defined to use Wireless InSite’s
GPU-accelerated X3D ray model, with ray-tracing defined to include propagation paths with up to six reflections and one diffraction, omitting transmissions to limit
calculations to the outdoors.
Simulations using the single antenna and the MIMO
antenna are then performed. Wireless InSite provides
user-requested outputs such as received power, propagation paths, time of arrival/departure, etc.
Figure 4: The Study Area Properties
window gives the user control overwhich propagation model to run andthe number of interactions.

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Example 1: 5G Massive MIMO Outdoor Communications Analysis
A typical result from the analysis of a single transmitting antenna is the power received
by a mobile device, either in the form of a coverage map over an area or displayed asa function of distance along a route. Figure 5 provides a field map showing the coverage in the vicinity of the base station. Figure 6 shows a plot of the power received by
the mobile device along its route (shown in red in views of the scene). As shown in thesecond figure, the power increases along the route as the mobile device approaches
the base station and falls off as it moves away, dropping dramatically when it turns thecorner and goes out of line-of-sight. Rapid fading in the received power along the routeis due to multipath interactions (reflections and diffractions) with buildings.

Figure 5: Field map showing the coverage over an
area of the scene.

Figure 6: Power received by mobile device along route frombase station with single antenna.

Another output that is relevant for both single antennas and MIMO systems is the complex impulse response, which is a measure of multipath. The complex impulse response
measures the power of arriving multipath signals as a function of either their time ofarrival or phase. Figure 7 shows the top 25 propagation paths between the base stationand mobile device at a point approximately 240 meters along its route. The complex impulse response at this location for the single antenna base station is shown in Figure 8.

Figure 7: Top 25 propagation paths from base station to mobile device at one point along its route.

Figure 8: Complex Impulse Response for mobile
device at position shown above.

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Example 1: 5G Massive MIMO Outdoor Communications Analysis
Wireless InSite MIMO allows users to write out H-matrix, Complex Impulse Response,and Delay Spread using a channel data results browser shown in Figure 9. The image shows settings for plotting received power as a function of receiver number for the
Rosslyn MIMO scenario for all combinations of transmitting and receiving antenna elements in the MIMO system. Figure 10 shows the received power at the mobile device
along its route, overlaying results for all transmitter elements.

Figure 9: The “Analyze Channel Data”
results browser was created for the Wireless
InSite MIMO module. This browser allows
users to output H-matrix, Complex Impulse
Response, Delay Spread and Received
Power.

Figure 10: Power received by mobile device
along route from all 128 transmitting elements of MIMO base station.

One of the key output results from a MIMO simulation is the complex channel matrix,or H-matrix. The H-matrix encodes the multipath propagation characteristics from everytransmitter element to every receiver element and is commonly used in beamforming,channel capacity, and other MIMO system-level analyses. Wireless InSite’s channel
data results browser allows a user to export the H-matrix to a .csv file, where additional
post-processing analysis can determine how channel multipath may impact the performance of the MIMO system.
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Example 2: mmWave Channel Modeling with Diffuse Scattering
Summary: The millimeter wave frequencies being planned for 5G systems posechallenges for channel modeling. At these frequencies, surface roughness impacts
wave propagation, causing scatter in non-specular directions that can have a large
effect on received signal strength and polarization. To accurately predict channel
characteristics for millimeter wave frequencies, propagation modeling must accountfor diffuse scattering effects. Wireless InSite’s diffuse scattering capability is based on
Degli-Esposti’s work.
The millimeter wave frequencies being planned for 5G systems pose challenges forchannel modeling. At these frequencies, surface roughness impacts wave propagation,
causing scatter in non-specular directions that can have a large effect on received signal strength and polarization. To accurately predict channel characteristics for millimeterwave frequencies, propagation modeling must account for diffuse scattering effects.
Wireless InSite’s diffuse scattering capability is based on Degli-Esposti’s work. It includes three models that provide alternative scattering patterns and account for partial
cross-polarization of the scattered fields. It also allows the user to optionally sum scattered contributions assuming coherent phase, so that phase effects over closely-spaced
antennas (e.g., MIMO) can be accounted for.
This example uses Wireless InSite’s diffuse scattering capability to perform simulationsof an indoor wireless network and compares to some measurements detailed in [1]. The
scenario, shown in Figure 1, is a portion of the 9th floor of an office building, which includes walls, pillars, windows, cubicle partitions, desks, and cabinets. The transmitter islocated at ceiling level in a large, open room. The receivers are located at several points
in the room and down the hallway.

Figure 1a

Figure 1b

Figure 1c

Figure 1d

Figure 1: Various positions of transmitting horn antenna aligned with various receiver points

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Example 2: mmWave Channel Modeling with Diffuse Scattering
We focus on the measurements done at 73.5 GHz in the large room nearest the transmitter. The transmitter is modeled using a horn antenna pattern with 15° beam width and
20 dBi gain that is rotated to point at each receiver location. The receivers are mounted
1.5m above the floor in several locations, both within line-of-sight, and beyond line-ofsite from the transmitter. Each receiver is modeled with both a vertical and horizontal
45° beamwidth Omnidirectional antenna.
The diffuse scattering effects of the floor, drywall walls, pillars, filing cabinets, cubicle
partitions, and desk supports are modeled using the directive model described in [4, 5,
6]. With this model, scattered energy is centered around the specular reflection angle
with no back scatter. The Scattering Factor, S, represents the fraction of the incident
electric field that is scattered diffusely. Valid values range from 0, no diffuse scattering,
to 1, all diffuse scattering. The cross-polarization fraction K-xpol represents the fractionof diffuse power that becomes cross-polarized relative to the incident ray’s polarization.
Valid values range from 0 to 0.5. Lastly, the parameter alpha controls the shape of theforward-scattering lobe. Valid values are integers between 1 and 10, with 10 being thenarrowest. Figure 2 shows how energy is scattered when using the directive diffuse
scattering model.

Figure 2: Directive diffuse scattering model

Recommended values for the Scattering Factor at 60 GHz range from 0.1 to 0.5 [2]. Table 1 shows the Scattering Factor used for each of the diffuse scatter enabled materials
in this example. Material properties for the drywall, concrete (floor, ceiling and pillars),
glass windows, and desktops are taken from [2]. The cubicle partitions are assumed to
be Fiber Reinforced Plastic (FRP) honeycomb core, and their properties are extrapolated from [3]. The filing cabinets and other metal structures are assumed to have handles,
beveled edges, and other structures that can contribute to scattering. Cross-pol fraction
and alpha were left at the Wireless InSite default values of 0.4 and 4 respectively.

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Example 2: mmWave Channel Modeling with Diffuse Scattering

Table 1: Scattering Factor for various building materials

Two study areas are created to allow for easy comparison of non-diffuse scatter results
and results with diffuse scattering. For the non-diffuse scattering interactions the settings are 3 reflections, 1 transmission and 1 diffraction. For the diffuse scattered paths,
we allow 1 reflection, no transmissions and 1 diffraction. The ability to allow interactions
along the diffuse scattering paths is unique to Wireless InSite.

Diffuse Scattering Properties window
in the X3D study area

To replicate the measurement procedures described in [1], it is necessary to align thetransmitter’s antenna with the intended receiver, and consider only the intended, paired
channels. This was done by creating seven transmitters at the location for transmitter 1,
each with the horn antenna rotated to align with the intended receiver location. Figure 3
shows the antenna alignment for receiver locations 1, 4 and 5.
The aligned pairs are most efficiently run by selecting and making active; for example,
TX1 to RX1, RX1 and RX1-xpol, running the simulation, then moving on to TX1 to RX2,
RX2, and RX2-xpol, etc. Running the simulations in this manner requires a little over aminute for each specular case, and about four minutes for each diffuse scatter simulation.

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Example 2: mmWave Channel Modeling with Diffuse Scattering

Figure 3a

Figure 3b

Figure 3c
Figure 3: Various positions of transmitting horn antenna aligned with various receiver points

Figures [4a] and [4b] show the top 100 paths between the matched TX- RX pair forthe receiver location 7. The red paths are the strongest, with weaker paths rendered ingreens and blues. The specular paths in Figure [4a] show the highest power paths are
typically within the main beam, with many lower powered reflections and transmissions
off and through the walls. Figure [4b] includes paths with diffuse scattering interactions.
One can see the wide spread of paths scattering backwards off the wall behind thereceiver. Other non-specular interactions off the pillars and partitions are also apparent.
The range in power of these paths is less dynamic than for the specular-only paths.

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Example 2: mmWave Channel Modeling with Diffuse Scattering

Figure 4a: Specular propagation paths to RX7

Figure 4b: Paths include diffuse scatter to RX7

Alternatively, the Complex Impulse Response (CIR) shows the power of each path asa function of its arrival time. Figures [5a] and [5b] show the CIR for RX7, for co-pol andcross-pol respectively. The specular-only results are plotted in blue, while the red results include diffuse scattering effects. The diffuse scatter results show more smearing,with arrival times occurring very close together, while the specular results are more
discrete and spread out.

Figure 5a: Co-polarized Complex Impulse Response for RX7 with and without diffuse scattering

Figure 5b: Cross-polarized Complex Impulse
Response for RX7 with and without diffuse scattering

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Example 2: mmWave Channel Modeling with Diffuse Scattering
The relevant path loss output files for comparison to measured data are listed in Table
2. A copy of these has been saved to the Aligned Antennas folder within the project. In
addition, the simulated path loss results have been collected into four files for simpler
plotting: Aligned_specular_CoPol.plt, Aligned_specular_XPol.plt, Aligned_DS_CoPol.
plt, and Aligned_DS_XPol.plt

Table 2Plot files for the measured results presented in [1] are located in the measurement folder. They can be plotted against the Wireless InSite data by importing their plots.
Figures [6a] and [6b] below show Wireless InSite predictions for path loss vs. distancefor the seven receiver locations for the co-polarized (V-V) case and the cross-polarized (V-H) case. The red line shows path loss for specular paths only, and the blue line
represents path loss with diffuse scattering.The measured results presented in [1] areshown in green. These plots show that diffuse scattering has a much greater impact on
the cross-polarized results and is critical for providing accurate predictions.

Figure 6a: Co-polarized TX/RX path loss - simulation with and without diffuse scattering compared tomeasured

Figure 6b: Cross-polarized TX/RX path loss - simulation with and without diffuse scattering compared tomeasured

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Example 2: mmWave Channel Modeling with Diffuse Scattering
References:[1] G. MacCartney, T.S. Rappaport, S. Sun, and S. Deng, “Indoor Office Wideband Millimeter-Wave Propagation Measurements and Channel Models at 28 and 73 GHz for Ultra-Dense 5G Wireless Networks,” IEEE Access, Vol. 3, Dec 7, 2015, pp. 2388 - 2424
[2] Effects of building materials and structures on radiowave propagation above about
100 MHz, Recommendation ITU-R P.2040-1, July 2015.
[3] A. Von Hippel, and W. B. Westphal, Tables of Dielectric Materials, Volume V, Final
Report, Laboratory for Insulation Research, Massachusetts Institute of Technology,
Cambridge, MA, April 1957.
[4] Degli-Esposti, V., F. Fuschini, E.M. Vitucci, and G. Falciasecca, “Measurement andModeling of Scattering from Buildings”, IEEE Transactions on Antennas and Propagation, Vol. 55, No. 1, January 2007, pp. 143-153.
[5] Degli-Esposti, V., “A Diffuse Scattering Model for Urban Propagation Prediction”,
IEEE Transactions on Antennas and Propagation, Vol. 49, No. 7, July 2001, pp. 11111113.
[6] Degli-Esposti, V., V.-M. Kolmonen, E.M. Vitucci, and P. Vainikainen, “Analysis andModeling on co- and Cross-Polarized Urban Radio Propagation for Dual-Polarized
MIMO Wireless Systems”, IEEE Transactions on Antennas and Propagation, Vol. 59,
No. 11, November 2011, pp. 4247-4256.
[7] J. Pascual-Garcia, et. al., “On the Importance of Diffuse Scattering Model Parameterization in Indoor Wireless Channels at mm-Wave Frequencies,” IEEE Access, February 8, 2016, © 2016 IEEE.
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Example 3: Simulation of MIMO Handset Antenna Array Performance
Summary: Performance of a 12-port handset antenna array operating in LTE bands
42/43 (3400-3800 MHz) and band 46 (5150-5925 MHz) is analyzed in XFdtd for varying
hand hold positions on the device. The results computed include S-parameters, Gain,
Efficiency and Envelope Correlation Coefficient.
A proposed design for a 12-port 5G massive MIMO mobile terminal was presented for operatingin sub-6 GHz range, specifically LTE band 42 (3400-3600 MHz), band 43 (3600-3800 MHz),and band 46 (5150-5925 MHz) [1]. This device was shown to have good performance for 8x8
MIMO operation in bands 42/43 and 6x6 MIMO operation in band 46 by using several different
antenna designs spaced around the edges of a mobile handset PCB. Here the performance of
the antenna is simulated in XFdtd first with the handset alone and then in various configurationsof either one or two hand grasping. The performance criteria evaluated are the return loss,
isolation between antennas, the efficiency, gain, and the envelope correlation coefficient (ECC)
which is an indicator of the diversity performance of antenna pairs.
Antenna Layout
The device is shown in Figure 1 where the 12 antenna elements are visible and labeled aroundthe edges of the PCB. At the top and bottom left are dual band inverted PI-shaped antennas
which operate in all bands under consideration. These antennas are sometimes labeled as IA in
the figures that follow and are numbered as Antennas 1 and 2. Along both sides of the antenna
are six longer inverted L-shaped open slot antennas (LA) for the lower (LTE 42/43) bands
and four shorter inverted L-shaped open slot antennas (SA) for LTE band 46. The antennas
alternate between LA and SA down the sides with three LA and two SA antennas per side. The
numbering of antennas is 3 through 7 on the right (numbers 3, 5, and 7 are LA) and 8 through12 on the left (with 8, 10, and 12 being LA). The dual band antennas are designed with a longer
and shorter arm for the lower and higher frequency bands. In Figure 2, Antenna 1 is shown with
conduction currents flowing at 3.6 GHz onto the longer arm and at 5.5 GHz on the shorter arm.
Two elements of the LA and SA antennas are shown in Figure 3 for comparison purposes.

Figure 1: A CAD view of thedevice in the XFdtd softwareinterface is shown where
the screen and all antennas
lie in the XY plane. The
antenna elements are numbered 1 through 12 aroundthe edge of the device.

Figure 2: At the left of the figure are shown the
conduction currents computed by XFdtd on thelonger portion of antenna element 1 at 3.6 GHz in
the LTE band 42/43 range. On the right are theconduction currents on the shorter arm of antenna
element 1 at 5.5 GHz in the LTE band 46 range.

315 S. Allen St., Suite 416 | State College, PA 16801 USA | +1.814.861.1299 phone |Figure 3: At the top of thisview of the XFdtd interface
is one of the longer arm
open slot (LA) antennas
for 3.6 GHz operation on
the edge of the device. At
the bottom is a shorter arm
open slot (SA) antenna for
5.5 GHz use.

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Example 3: Simulation of MIMO Handset Antenna Array Performance
Simulation in Free SpaceAs one measure of performance, the S-parameters of various antennas will be considered toshow the return loss and isolation of the antennas. In Figure 4, the return loss for antennas 1
and 2 (the top and bottom IA antennas) is shown along with the S21 between them. As canbe seen, the antennas have good return loss values in the lower LTE 42/43 bands and thehigher LTE 46 band. S21 remains low over both bands, which is expected since the antennas
are on extreme ends of the PCB. The return loss of all the LA antennas in LTE bands 42/43
is shown in Figure 5. Figure 6 shows the return loss of the SA antennas in band 46 alongwith the isolation between adjacent SA antennas and between adjacent SA and IA antennas.
The isolation between adjacent LA antennas (and between adjacent LA and IA antennas) forthe lower frequency bands is shown in Figure 7. In all cases the results are within the design
tolerances of return loss under -6 dB (3:1 VSWR) and isolation -11 dB or better.

Figure 4: The S-parameters for antennas 1 and 2 areshown over all bands of operation of the device. There
are two distinct operating regions: around 3.6 GHz andaround 5.5 GHz. For both, the return loss is well below
required levels while the isolation determined by S21 isdown around -18 dB.

Figure 6: The return loss and isolation of select elements
is shown for the higher band (LTE 46) with good results.

Figure 5: The return loss for all antennas in the lower band
(LTE 42/43) are shown and have very good results.

Figure 7: The isolation between select elements in LTE
bands 42/43 are shown.

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Example 3: Simulation of MIMO Handset Antenna Array Performance
The antenna performance is determined by viewing the efficiency and gain of the various
elements over their bands of operation. In Figure 8 the efficiency of the LTE Band 42/43
antennas is shown and is reasonable for the IA antennas (1 and 2) and moderate for the LA
antennas (3, 5, 7, 8, 10, and 12). In the LTE 46 band, the antenna performance is better, withefficiencies above 50% as shown in Figure 9. Gain is plotted in the XY plane (the plane of thedevice screen) for various combinations of antennas. In each case, the antenna gain patterns
are designed to radiate away from the center of the device. In Figure 10, the gain for the IA
antennas at LTE bands 42/43 are shown and can be seen to have a maximum gain at a 45
degree angle from the antenna which is toward the corner of the device screen. In Figure 11,
the gain for the LA antennas in the LTE 42/43 band on the right side of the phone is shown and,in each case, the maximum gain is along the X axis, away from the screen center. In Figure 12,
the gain for the SA antennas on the right side of the phone in LTE band 46 is shown with similar
characteristics.

Figure 8: The efficiencies for antennas in the LTE band
42/43 are shown. The two IA antennas at the ends of thephone (1 and 2) show higher efficiency over 60% while theLA antennas on the sides of the phone have lower efficiencies between 35 and 50%.

Figure 10: The gain patterns in the XY plane for the two IA
antennas show peak gains toward the respective cornersof the device.

Figure 9: The efficiencies for all antennas in the higher
LTE 46 band are above about 50% and show good performance.

Figure 11: The LA antennas have peak gain toward theouter edge of the device. In this case, the antennas on theright side of the device are shown with peak gain in the +X
direction.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Figure 12: The SA antennas have peak gain toward the outer edge of thedevice similar to the LA antennas. Here antennas 4 and 6 are shown withpeak gain in the +X direction.

To evaluate the diversity performance of antenna pairs, an important measure beyond theisolation is the envelope correction coefficient (ECC). This measure is useful for determining
how well a device will work for diversity and multiplexing applications where it is important that
individual antennas perform independently from others. The basic criterion for ECC is a valueless than 0.5. The ECC for various antenna pairs in the LTE 42/43 band and LTE 46 band
are compared in Figures 13 and 14, respectively. All plots are well below the 0.5 limit with the
maximum correlation being between Antennas 8 and 10 in the LTE 42/43 band at about 0.15.

Figure 13: The Envelope Correlation Coefficient (ECC)
for the LTE band 42/43 antennas is quite good with a peakvalue of 0.15.

Figure 14: The Envelope Correlation Coefficient (ECC)
for the LTE band 46 antennas is very good with no twoantennas higher than 0.05.

Simulation with Hands
To further test the phone operation in more realistic conditions, hand models are introduced
in both single hand and two hand configurations, as shown in Figure 15. The positions arefor left and right hand holds, two hands at the side mode, and two hands with thumbs typing
configuration. Depending on the hand position, certain antennas will be covered and theperformance will be affected. However, since there are multiple antennas on both sides of thedevice, in most situations there are still several antennas available with good performance.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Figure 15: Shown are the four positions of the Poseable Hand model used in the study,
which were positioned using controls within XFdtd. Clockwise from top left: left handhold, two hands at sides, two hands typing, and right hand hold.

The S-parameter performance remains good for all hand positions, although in some cases
the return loss is significantly affected. For example, in Figure 16 the return loss for the LTE
band 42/43 antennas is shown for the left and right hold positions and there is are noticeable
difference versus the phone alone return loss shown in Figure 5. Similarly, the SA antennas
operating in the LTE band 46 are also changed by the left and right hand hold positions
as shown in Figure 17. For the two hand hold positions, there is less disruption to the
S-parameters for the same antennas since the hands are not directly covering the antennas, as
shown in Figures 18 and 19.

Figure 16: With the left and right hands holding thephone, the return loss results for the LTE band 42/43
antennas are changed but still good enough.

Figure 17: With the left and right hands holding the phone,
the return loss and isolation remain acceptable for the LTE
band 46 antennas.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Figure 18: In the two hand configurations, the returnloss in the LTE 42/43 band is only slightly changed andremains acceptable.

Figure 19: At the higher LTE band 46, the two hand configurations have only slight impacts on the return loss andisolation.

As can be expected, the efficiency is greatly affected by the presence of the hands. The
patterns are disrupted and power is lost in the hand tissue resulting in worse performance.
An extreme example is shown in Figure 20 for the case of the two hands at the sides in LTE
bands 42/43 where the peak efficiency is only around 35%. Other cases studied had better
performance.

Figure 20: Efficiency is greatly
affected by the hands on thephone. Here the efficiency for
LTE bands 42/43 for the two
hands at sides case is shownwith no antenna above 35%
efficiency.

The gain patterns are understandably impacted by the hand positions on the phone. Figure 21
shows the three-dimensional patterns of the six LA antennas at LTE bands 42/43 in the caseof the phone alone. When the left hand is holding the phone, as shown in Figure 22, some ofthe patterns, particularly antenna 10 by the thumb, are greatly impacted and show much lessgain. For the higher frequency LTE band 46 SA antennas on the sides, there are fairly uniform
patterns when the device is alone, such as in Figure 23. With the hands at the sides and thefinger next to the two bottom antennas, as in Figure 24, the patterns on the bottom antennas(4 and 6) are reduced compared to the top antennas (9 and 11). For the two hand typing
configuration and the same antennas, the top antennas show an increase in gain due to thepresence of the thumbs in the middle of the screen as shown in Figure 25. In summary, thepeak gains for all configurations are detailed in Tables 1 and 2.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Figure 21: Three-dimensional gain patterns of the six
LA antennas are shown at 3.6 GHz (LTE bands 42/43)
for the case of the device alone in free space. Eachantenna is radiating a pattern with peak gain to theouter edges of the device as desired.

Figure 22: With the left hand holding the phone in LTE
band 42/43, the gain patterns are modified and the center
pattern under the thumb (antenna 10) is greatly reduced ingain.

Figure 23: In LTE band 46, the three-dimensional gain
patterns are shown with peak gain toward the outsideof the device.

Figure 24: In the two hands at side position in LTE band46, the gain patterns at the bottom of the phone (antennas
4 and 6) are disturbed by the presence of the fingers andare reduced compared to the upper patterns.

Figure 25: In the two hand typing configuration in LTE band 46 the upper patterns (antennas 9
and 11) are enhanced by the impact of the thumbs with give a reflector effect.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Table 1: The peak gains for each antenna at 3.6 GHz (LTE bands 42/43) are shown for the five configurations.

Table 2: The peak gains for each antenna at 5.5 GhHz (LTE band 46) are shown for the five configurations.

As can be expected, the ECC results are also affected by the presence of the hands. In allcases though, the ECC remains well below the 0.5 threshold as desired. The worst-case ECC
results are for the left and right hand hold positions in the LTE band 42/43 antennas where the
ECC can reach as high as 0.4 as shown in Figures 26 and 27. All cases in LTE band 46 remain
with an ECC below 0.2.

Figure 26: The worst-case ECC is shown for the right hand hold case at LTE bands 42/43
where the correlation between antennas 3 and 5 reaches as high as 0.4. This is still below
the threshold of 0.5.

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Example 3: Simulation of MIMO Handset Antenna Array Performance

Figure 27: In the left hand hold position at LTE bands 42/43, the ECC is highest for antennas 8 and 10 with a peak level around 0.33.

The 12-port antenna can be seen to deliver very good performance with a variety of use
configurations. While some hand placements degrade the performance of antennas that arecovered, there are other antennas available to maintain operation.

Reference:[1] Yixin Li, Chow-Yen-Desmond Sim, Yong Luo, and Guangli Yang, “12-Port 5G Massive
MIMO Antenna Array in Sub-6GHz Mobile Handset for LTE Bands 42/43/46 Applications,” IEEE Access vol. 6, pp. 344-354, 2018.
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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G
Summary: Series-fed patch elements forming an array are simulated to demonstrateantenna performance and beamforming including S-parameters, gain, and effective
isotropic radiated power (EIRP) at 28 GHz. Beam steering is performed in one planeby adjusting the phasing at the input ports to each of eight elements.
Part of the discussion for next generation wireless communication is the ability to
rapidly steer beams from antenna arrays at higher frequencies. In this example, aproposed antenna [1] is comprised of eight series-fed patch elements, each containing
eight patches that are connected in such a way to produce both standing wave andtraveling wave behavior. The array produces a single beam which may be swept in
the horizontal axis of the array by varying the phasing of the signals at the inputs of theelements.
The array is evaluated for standard performance metrics such as S-parameters andgain as well as for effective isotropic radiated power (EIRP). EIRP is effectively the
input power supplied to an antenna multiplied by the gain of that antenna. It indicates
the total power that an isotropic antenna would have to radiate to match the signal
strength of the main beam of the evaluated antenna. Simulations in this paper are
performed using XFdtd® EM Simulation Software.
The final array is built from eight separate 1x8 series-fed patch elements such as thatshown in Figure 1. The characteristics of a single element will be examined first before
discussing the array. Each patch is 3.539 mm x 3.539 mm and is separated 3.539 mm.
The last patch in the line has a notch cutout of 0.6269 mm x 2.727 mm in the connecting
side. The microstrip lines connecting the patches are 0.494 mm in width. The initialfeed line is 2.215 mm long and 0.72 mm wide. The total length of the element is 55.3
mm. The antenna is on a substrate with a dielectric constant of 2.2, a loss tangent of
0.0009 and a thickness of 0.254 mm.

Figure 1: CAD representation of a single 1x8 patch element viewed from the top.

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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G
The geometry is gridded into an FDTD mesh using the PrOGrid feature in XFdtd.
Each element of the geometry has Automatic Fixed Points enabled to perfectly alignthe corners of the structures with the FDTD grid. The minimum feature size for Good
Conductors is set to be the width of the microstrip line between patches and that isdefined to have at least eight FDTD cells across it. The dielectric substrate is defined
to have at least five FDTD cells of thickness using the Poor Conductor minimum featuresize setting. The overall number of cells per wavelength is set to 60 to ensure goodresults.
The input for the element is a voltage source
which is initially excited by a broadband
signal covering from 26 to 30 GHz for
S-parameter results. Following a simulation
for computing the S-parameters, the returnloss is found to have a weak response due
to a poor match. This is corrected by adding
a matching circuit consisting of a series
inductor and parallel capacitor (0.22 nH and
0.09 pF). The matched input return loss
produces a null near 28 GHz (Figure 2). The
far field gain pattern for the single element
Figure 2: The Return Loss for the 1x8 element is slightly(Figure 3) indicates a strong central lobe
out of tune when fed with a 50-ohm source. By adding
normal to the patches with peak gain of 16.77 a simple LC matching circuit, the device is tuned to 28
dBi. The peak side lobe is seen to be down GHz.
about 13 dBi from the main lobe as shown in
Figure 4.

Figure 3: The far-field gain pattern of the1x8 element has a strong central beam that is focused in
one dimension and circular in other. The peak gain
is just under 17 dBi.

Figure 4: In the XZ dimension (E-plane) the antenna
pattern has main beam with gain of 16.77 dBi and a peakside lobe level of 3.73 dBi.

To create the array, eight of the single 1x8 elements are spaced 5.352 mm apart toform an array that measures 55.3 mm x 41 mm as shown in Figure 5. To generatebroadband S-parameter data, a pulse excitation is applied to each of the eight input
ports.

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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G
A matching circuit similar to that used for the single element is applied to all input portsof the array. The return loss for all eight ports is seen to be very similar as shown in
Figure 6, while the isolation between adjacent ports is below -15 dB for each (Figure 7).

Figure 5: Shown is a CAD representation of the combination of eight of the 1x8 elements into an array. The
elements are spaced 5.352 mm apart, center to center.

Figure 7: The isolation between adjacent ports isshown to be less the -15 dB for all possible combinations.

Figure 6: The Return Loss of each port is shown
be tuned for 28 GHz after the addition of simple LC
matching circuits to each feed.

Figure 8: The gain pattern for the array when all inputs
are fed in phase results in a strong central beam withgain of 24 dBi.

Depending on the phasing of the signals at each of the input ports, a number of different
beams may be defined. When all ports are fed in phase, the beam normal to the arrayplane is formed with the maximum gain of 24 dBi (Figure 8). Due to the nature of thisgeometry the beams may only be steered in one plane, parallel to the line of input ports.
To adjust the phasing, the Butler Matrix equation is used to compute the inter-element
phase difference. It is defined as:
αi = [(2i – 1) – M]/M * π
where αi is the phase difference between elements, i is the beam number or input port
number of the Butler Matrix, and M is the number of ports. In this case, the phasing forbeams 1-8 is -157.5 deg, -112.5 deg, -67.5 deg, -22.5 deg, 22.5 deg, 67.5 deg, 112.5
deg, and 157.5 deg.

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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G
The phase shift is applied across the input ports, so the first port will have a shift of 0deg, the second -157.5 deg, the third -315 deg, etc. for the case of beam 1. These
phase shifts are applied with a sinusoidal input at 28 GHz. The resulting beams in theYZ (E-plane) are shown in Figure 9 as a series of line plots with main lobe directionsof +/- 55 degrees, +/- 37 degrees, +/- 21.5 degrees, and +/- 7 degrees. In three
dimensions, beams 1 through 4 may be seen in Figures 10 - 13. All eight beams areshown in one three-dimensional image in Figure 14.

Figure 9: After applying Butler Matrix phase shifts
to each port, a directed beam is formed. Shown areeight possible beams.

Figure 10: The gain pattern formed from the Butler Matrix phasing for beam #1 (-157.5 degrees per port) has abroad beam with maximum gain at 55 degrees.

Figure 11: The gain pattern formed from the Butler
Matrix phasing for beam #2 (-112.5 degrees perport) has a more focused beam with maximum gain
at 37 degrees. The white arrow indicates the direction of the peak gain.

Figure 12: The gain pattern formed from the Butler
Matrix phasing for beam #3 (-67.5 degrees per port) has
maximum gain at 21.5 degrees.

Figure 13: The gain pattern formed from the Butler
Matrix phasing for beam #4 (-22.5 degrees per port)
has maximum gain at 7 degrees.

Figure 14: Shown is a side view of the eight beams
produced by the Butler Matrix phasing equation. Eachbeam represents a separate simulation.

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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G
This array has been shown to form eight beams with the Butler Matrix phasing;
however, a further analysis of the total gain over all possible phasing combinations is
possible by computing the Cumulative Distribution Function of the Effective Isotropic
Radiated Power. This plot shows the fractional area of the three-dimensional far-field
sphere that the array covers for a given amount of input power. For this array, the
CDF of EIRP plot (Figure 15) shows that for an input power of 23 dBmW, the array has
positive gain over about 27.4% of the far-zone sphere. This represents the scanningarea where the beams have positive gain, which is limited by the ground plane under
the array that blocks 50% of the radiation volume. The upper hemisphere of the farzone region is scanned by the beams in only one axis due to the design of the array,so therefore the scanned region is relatively limited. The plot also shows that the peak
EIRP is about 46.4 dBmW.

Figure 15: The plot of the cumulative distributionfunction of the EIRP shows that for 23 dBmW of input
power, about (1 - .726) or 27.4% of the three-dimensional far field sphere will have positive gain.

Figure 16: A Rotman Lens device suitable foruse as the front end of the array is shown in
Remcom’s RLD software.

As an alternative to using the Butler Matrix equations and adjusting the phases inthe software, it is also possible to use a true time delay device such as a Rotman
Lens in the simulation to form the beams with this array. Using Remcom’s Rotman
Lens Designer® (RLD), a suitable lens (shown in Figure 16 in the RLD software) was
designed to meet the criteria of the array. After exporting the CAD file of the lens from
RLD and importing it into XFdtd, a complete geometry is made with eight input ports
and four sidewall dummy ports (Figure 17). In the simulations the beams are createdby feeding a single port at a time rather than adjusting the phasing over each port.
Following eight simulations, each with a different port active, the eight beams formed by
the array are visible in Figure 18. The coverage is similar to the Butler Matrix phasing
shown previously in Figure 14.

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Example 4: EM Simulation of 28 GHz Series-Fed Patch Antenna Array for 5G

Figure 17: The resulting geometry of the array with
the Rotman Lens beamforming stage are shown asa three-dimensional CAD file in XFdtd.

Figure 18: The eight possible beam patterns fromthe array generated by activating one of the ports of
the Rotman Lens at a time are shown.

Reference:[1] M. K. Ishfaq, T. A. Rahman, Y. Yamada, and K. Sakakibara, “8x8 Phased Series Fed Patch
Antenna Array at 28 GHz for 5G Mobile Base Station Antennas,” 2017 IEEE-APS Topical
Conference on Antennas and Propagation in Wireless Communications (APWC), pp. 160-162,
2017.
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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area
Summary: This example demonstrates how a custom beamforming table can be used
to model downlink data rates from three MIMO base stations for 5G New Radio in asection of Boston.
Wireless InSite’s Communication Systems Analysis module is a set of post-processing
routines that builds on the X3D model’s high-fidelity MIMO calculations to predict
system throughput and bit error rate. This example utilizes these routines to analyze
throughput between three small-cell base stations employing Full Dimension (FD) MIMO
beamforming to User Equipment (UE) moving along a route, using 5G New Radio (NR)in a dense urban environment.

Project Description
The building and foliage geometry was imported from a high resolution shapefile ofgeometry for a section of downtown Boston, MA. The imported city model, foliage,
and the terrain contain a total of 6,452 faces. The buildings and terrain are assigned
concrete material properties with conductivity of 0.484 S/m and permittivity of 5.31 takenfrom ITU data for concrete at 28 GHz [1], [2], while foliage is modeled using a foliage
penetration loss model by Weissberger [3].

Figure 1: Isometric and overhead view of projectgeometry in Wireless InSite.

Each transmitter represents a base station and uses an 8x8 dual polarized patch arrayconsisting of a total of 128 elements. Cross-polarized isotropic antennas are used
at receiver locations along the route. Wireless InSite’s GPU-accelerated X3D model
was used to provide high fidelity propagation results which include reflections from thebuildings and terrain and diffractions from building edges. The combination of highresolution geometry and high fidelity propagation modeling is necessary to achieve
accurate results at 5G NR frequencies. The propagation model calculates the H-matrix
from each element of the three transmitters to the receiver locations (128 x 2 = 256
entries per tx-rx pair).

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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area

Communication Systems Analyzer
Wireless InSite’s Communication Systems Analyzer provides capabilities for predicting
throughput, bit error rate, and other outputs from multiple transmitters by postprocessing the high-fidelity H-matrix results from the propagation simulation. Within acommunication system, users can set input parameters for:
•
•
•
•
•
•
•

Selecting transmitter and receiver sets to be included in the analysis
Designating transmitters as base stations or interference sources
Setting uniform noise and interference levels
Optionally including interference between base stations
Selecting BER or throughput analysis
Selecting transmitter beamforming options
Setting diversity options for MIMO receiving antennas

In this example, a communication system is defined, specifying analysis of throughput
between three base stations and a UE traveling along a mobile route. The
communication system properties window defines a custom beamforming table for thebase stations, the use of selection combining by the UE, and a custom throughput tablebased on 3GPP specifications to be used in calculating the throughput for the link.

Figure 2: Communication system properties window for
calculating throughput for a MIMO system.

Custom Beamforming with a Precoding TableThe user-defined precoding table is a text file containing antenna element weightings
for the transmitting MIMO antenna. Each record in the file represents a beam that theantenna elements are capable of producing. This example uses a table that describes
66 antenna beams that allow the transmitter to scan horizontally from -50 to 50 degrees
and vertically from -10 to +40 degrees in 10 degree increments, by varying the phase ofeach antenna element. Base stations will select the antenna beam that maximizes thepower at the receiver for each position of the UE.

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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area
Custom Throughput Analysis for a 5G NR Device
The throughput for the 5G NR device at the receiver also uses a custom file, mapping
SINR at the device to an achievable downlink data rate. Throughput values for thistable were calculated using the equation published in 3GPP TS 38.306 v15.2, whichcalculates data rates for 5G NR for each potential modulation and coding scheme(MCS) and takes into account overhead for signaling and control and differences
between uplink and downlink transmission [4] [5]. Estimates were for a single
component carrier with a 100 MHz bandwidth and assumed similar relationships
between the achievable MCS and the SINR as has been observed for LTE.

Results
Receiver Diversity
Three communication systems were created to analyze the effects of selection
combining, equal gain combining, and maximal ratio combining for the 5G NR
user equipment (UE). Results in Figures 3 and 4 show that in this scenario andconfiguration, there is not a substantive difference between the three diversity methods
for SINR or throughput. The remainder of this example will use selection combining for
the UE and custom beamforming at the base stations.

Figure 3: SINR comparison of receiver
diversity methods for base station 1.

Figure 4: Throughput comparison of receiver
diversity methods for base station 1.

SINR and Throughput for Custom Beamforming and Selection Combining
SINR and throughput results are provided for each individual base station and for all
three base stations working together. In this example, the 5G NR will establish a linkwith the base station that provides the highest SINR value. The red line in Figures
5 and 6 represents the overall performance of the three base stations to the UE,
according to the base station that achieves the highest SINR at each point along theroute. The blue line represents the contribution of base station 1 to the SINR andthroughput. In regions where the blue and red lines overlap, the receiver establishes
a link with base station 1. Contributions from the other two base stations are alsoplottable but were omitted for clarity of the graphs.

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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area

Figure 5: Total SINR (red) and SINR fromBase Station 1 to UE (blue).

Figure 6: Total throughput (red) andthroughput from Base Station 1 to UE (blue).

Evaluation of the Beamforming Table vs. Maximum Ratio Transmission
Maximum Ratio Transmission beamforming calculates an optimal beam to each
receiver location. In Figures 7 and 8, the SINR and throughput results using a custom
beamforming table (blue) are compared against MRT (red). In most locations thepredefined beams from the custom beamforming table are able to achieve nearly thesame SINR and throughput as MRT (a more optimal, adaptive technique), but there aresections where adding more targeted beams would improve the results. Between 225
and 350 m along the route, the SINR values are too low to allow adequate throughput
values with both beamforming methods. To improve coverage in this location, it would
be necessary to add an additional base station.

Figure 7: Comparison of SINR for custom(blue) and MRT (red) beamforming.

Figure 8: Comparison of throughput along the
UE route from all three base stations for MRT(red) and custom beamforming (blue).

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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area

Figure 9: SINR coverage for custom
beamforming and selection combining.

Figure 10: Throughput coverage for custom
beamforming and selection combining.

Conclusion
Wireless InSite’s Communication Systems Analyzer can be used to calculate
throughput, SINR and other system level quantities for realistic MIMO systems,
including both transmit beamforming and receiver diversity.
This example demonstrates how a custom beamforming table can be used to modelthe dynamic behavior of three MIMO base stations in a section of Boston. A custom
throughput table was used to model downlink data rates for 5G New Radio. Wireless
InSite’s X3D high fidelity ray tracing and MIMO capabilities allowed for efficient
calculation of SINR and throughput.
This example also demonstrates how built-in theoretical beamforming and receiver
diversity options allow quick comparison of the throughput coverage provided by custom
techniques.
Remcom’s Communication Systems Analyzer provides network planners and equipment
designers with the tools needed to evaluate base station location and estimate thethroughput in realistic scenarios. With the ability to incorporate custom beamforming
and throughput rates, users have the flexibility to design and assess new equipment
and network configurations.

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Example 5: Throughput of a 5G New Radio FD-MIMO System in an Urban Area

References:[1] “Electrical characteristics of the surface of the earth,” Recommendation ITU-R P.5273, 1992.
[2] “Effects of building materials and structures on radiowave propagation above about
100 MHz,” Recommendation ITU-R P.2040-1, pp. 22–23, July 2015.
[3] M. Weissberger, “An Initial Critical Summary of Models for Predicting the Attenuationof Radio Waves by Trees,” 1982, Electromagnetic compatibility analysis center,
Annapolis, Maryland. Final report. https://apps.dtic.mil/dtic/tr/fulltext/u2/a118343.pdf[4] 3GPP TS 38.306 V15.2.0 [10] for 5G NR was used to calculate peak throughput
based on the modulation order and coding rate.
[5] G. Skidmore, “Using Modeling and Simulation to Assess Challenges and Solutionsfor 5G Fixed Wireless Access.”

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation
Summary: A proposed smartphone design that includes a 4G antenna operating
at 860 MHz and a 5G array at 28 GHz is analyzed in XFdtd to determine operating
characteristics and any mutual coupling. A brief study of configurations is performed tofind the best positioning for each antenna.
Introduction
This example uses XFdtd® EM Simulation Software to analyze the performance andinteraction of two antenna systems operating at 4G (860 MHz) and 5G (28 GHz) inclose proximity in a smartphone design. The 4G antenna is intended to produce a
broad pattern for wide coverage while the 5G array should produce narrow beams
that can be steered by varying the phasing between elements. The 4G antenna is aninverted-L design and is located at the top of the phone. The 5G antenna array consistsof four Yagi-Uda elements that are near the 4G antenna but offset by a conducting
block. The antenna designs used in this example are from the paper [1].
Device Geometry
The example’s geometry is based on a simplified smartphone PCB. A 150x70x1 mm
sheet of copper is used as the ground plane for the system with the antenna structures
mounted on one end, as shown in Figure 1. The 4G antenna is placed directly over the
ground plane in an inverted-L shape that has a 1x1 mm cross section and lengths of 6
mm on the short leg and 70 mm on the long leg. It is fed by a voltage source between
the ground plane and the end of the short leg. The 5G array is comprised of four
identical Yagi-Uda elements shown in Figure 2. The elements are mounted on a 0.3
mm thick substrate (dielectric constant = 3.6, loss tangent = 0.006) which is separated
from the ground plane by a conducting block 2.2 mm thick. There is also a horizontal
element of the Yagi-Uda antenna on the back side of the substrate which is visible in
Figure 3. Each 5G element is fed by a Nodal Waveguide source.

Figure 1: The smartphone design is shown in athree-dimensional CAD representation with the
4G antenna and 5G array structures visible at theleft on one end of the large ground plane.

Figure 2: A CAD view of the 5G array showing
the four Yagi-Uda elements on a dielectric substrate with the 4G inverted L antenna behind.

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation

Figure 3: An underside view of the antenna array shows the separation of the 4G
and 5G elements and the back side of the substrate.

The structure is meshed using the PrOGrid Project Optimized Gridding® feature of
XFdtd with a base setting of 30 FDTD cells per wavelength. Minimum feature sizes
are defined to ensure the substrate has five FDTD cells of thickness and the smallest
conductor features are five cells across. Automatic fixed points are also applied to allparts to ensure grid lines are placed on all the edges of the geometry.
Results
The 4G antenna is fed by a broadband source which is initially represented as a 50-ohm
voltage source attached between the ground plane and the antenna base. The antenna
is not a good match to the 50-ohm source resulting in poor return loss. After an initial
simulation to determine the impedance of the antenna load, a matching circuit canbe defined to improve the antenna performance. In this case, a lowpass-PI matching
network consisting of two capacitors and an inductor is selected for use and component
values are determined using analytic matching circuit calculators. The matching
network components are defined in a SPICE netlist format and then imported into XFdtd
as a sub-circuit as shown in Figure 4. This sub-circuit can then be added to an XFdtd
voltage source as a matching circuit (Figure 5). During the execution of the program, aco-simulation of the FDTD full wave calculation and a circuit solver takes place for eachtime step at the cell edge containing the matching circuit. A normal full wave FDTD
simulation takes place for the remainder of the calculation space. The resulting returnloss plots for the unmatched and matched 4G antenna are shown in Figure 6 and themarked improvement from the matching circuit is visible. The gain of the 4G antennais shown in Figure 7 and can be seen to be provide broad coverage with a peak gain of
2.7 dBi.

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation

Figure 4: The menu for the netlist of the matchingcircuit used in the 4G antenna is shown. The netlist contains two capacitors and one inductor.

Figure 5: Shown is the voltage source menu
in XFdtd for the port with the matching circuitthat is used for the 4G antenna.

Figure 6: Return loss comparison for the unmatched and matched 4G antenna.

Figure 7: The 3D gain pattern of the 4G antenna shows fairly uniform coverage around thedevice and a peak gain of 2.7 dBi.

The orientation of the 5G antenna array relative to the 4G antenna is varied todetermine the impact of the positioning on the performance of each antenna. Three
configurations are considered and are shown in Figure 8: shift the 5G array so the topsof the 5G antennas are 2 mm below the top of the 4G antenna, keep the tops of the 5G
and 4G antennas at the same level, or shift the 5G array up by 2 mm. In these initialcases, the phase shift between the 5G array elements is kept at zero. The impact of theoffset is displayed in Figure 9 and Figure 10 where it can be seen that the offset of the
5G array down by 2 mm has a noticeable impact on the gain pattern in both the XY andYZ planes. Moving the 5G array up by 2 mm produces a pattern similar to when the 4G
antenna is completely removed from the geometry. When the tops of the 4G and 5G
antennas are aligned, there is a slight impact to the 5G antenna pattern. In all cases,
the 4G antenna pattern is only slightly impacted, as can be seen in Figure 11.

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation

Figure 8: Three different configurations of the 5G array relative to the 4G antenna are considered (left to right):
an offset of the top of the 5G antennas 2 mm below the 4G antenna, even alignment between the top of the 5G
antennas and the 4G antenna, and an offset of the top of the 5G antennas of 2mm above the 4G antenna.

Figure 9: Plots of the gain pattern in the XY planeof the 5G array with zero phase shift between
elements are shown. It can be seen that with the
5G array offset by 2 mm above the 4G antenna,
the pattern is similar to when the 4G antenna isremoved from the geometry. Similar patterns arefound when the tops of the 4G and 5G antennas
are aligned. When the 5G antennas are offset 2
mm below the 4G antenna, the 5G pattern is significantly affected.

Figure 10: Plots of the gain pattern in the YZ planeof the 5G array with zero phase shift between
elements show similar results for the 2mm above
offset to the case of the 4G antenna removed.
There is some impact on the 5G array patternwhen the tops of the antennas are aligned and alarger impact when the 5G antennas are shifted 2
mm below the 4G antenna.

Figure 11: The gain pattern of the 4G antenna in the XY plane isonly slightly impacted by the movement of the 5G antenna array.

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation
Next, the 5G antenna array is fed by a 28 GHz sinusoidal source with equal amplitude
and a variable phase shift across the elements for the configuration where the 5G
array is positioned 2 mm above the 4G antenna. With the array fed with all elements inphase, the pattern produced has a peak gain of about 12.5 dBi with a 3 dB beamwidthof 17 degrees and a peak sidelobe level below 0 dBi (Figure 12). After applying a phaseshift of 120 degrees between each element of the 5G array, the beam is scanned 24.5
degrees to one side as shown in Figure 13. Applying phase shifts of -180, -120, -60, 0,
60, and 120 degrees to the feeds produces a series of beams from the array, shown in
Figure 14 where all beams are plotted at the same time.

Figure 12: The gain pattern of the 5G array with zero phase shift between elementsshows a beam with peak gain of 12.5 dBi and a beamwidth of 17 degrees in the XY
plane.

Figure 13: The gain pattern of the 5G array with a 120 degree phase shift between
elements shows a beam tilted 24.5 degrees in the XY plane.

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Example 6: Combined 5G/4G Antenna Analysis in a Smartphone using EM Simulation

Figure 14: Six possible beams of the 5G array are shown in 3D and 2D gain
patterns for the cases of a phase shift between elements of 180, 120, 60, 0, -60,
and -120 degrees.

Conclusion
A simplified smartphone device with a 4G antenna combined with a 5G array at 28 GHz
is evaluated for different configurations of the 5G array relative to the 4G antenna. A
best-case configuration with the 5G antenna offset 2 mm vertically relative to the 4G
antenna was found with results that indicate there is nearly no interaction between the
4G and 5G elements.
Reference:[1] T. Yamagajo, Y. Koga, M. Kai, T. Tonooka, H. Sumi, and M. Hoshino, “A Nobel 4G
and 5G Antenna Solution for Future Smartphones,” IEEE Antennas and Propagation
Society Int. Symp., pp. 1785-1786, 2018.
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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array
Summary: An 8x8 planar antenna array creates narrow beams capable of scanninglarge sectors in front of the antenna. This example focuses on displaying typicalsimulation results for beams and possible plots of coverage from the full array andcombinations of sub-arrays.
Antenna Setup
The antenna geometry consists of a 52.5 x 52.5 mm sheet of
dielectric material (dielectric constant = 2.2, loss tangent 0.0009,
thickness 0.254 mm) that is backed by a conducting ground plane
and topped with 64 patch elements in an 8x8 configuration. Each
patch is 3.4025 mm square and is spaced a half-wavelength at 28
GHz (approximately 5.36 mm) apart. A view of the patch layout on
the dielectric sheet is visible in Figure 1. The feed point for eachantenna is offset from the center of the patch by 0.75 mm in thehorizontal direction as this was found to be the location of the best
return loss result. A 28 GHz sinusoidal waveform is used at eachpatch with an adjustable phase offset that is determined based on
the desired direction of the main beam. A widely-used formula for
determining the phasing of the elements for a beam focused at a
direction of θd, φd isFigure 1: A top view of theantenna geometry showing the layout of the 8x8array of patches.

Wn = exp{-j(2π/λ) sin(θd)[ xn cos(φd) + yn sin(φd)]}where xn and yn are the locations (in meters) of the feeds at each patch referenced tothe initial patch in the lower left corner of the array, and wn is the phase shift for theelement located at (xn, yn). In XFdtd, these phases were assigned to each feed element
through the use of parameters as shown in Figure 2, where the phase shift is defined bya parameter name.

Figure 2: An example of one of the source definitions
for the patch feeds, showing the phase shift set asa variable which may be adjusted depending on thedesired beam direction.

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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array
Results
Simulations were performed on the array to determine the gain pattern possible fordifferent phasing conditions. An initial test set all phases equal so all elements areradiating toward the (0°, 0°) direction. This resulted in a maximum gain of just over
23 dBi and a 3 dB beamwidth of just over 12 degrees as shown in a line plot throughone of the principal planes in Figure 3. The same pattern is shown over the CAD
drawing of the antenna in three dimensions in Figure 4. With the phases adjusted todirect a beam toward (20°, 90°) the result is a slight drop in peak gain to 22.9 dBi anda broadening of the beamwidth to about 13.2 degrees, shown in three dimensions inFigure 5. Sweeping the beam to a corner of the array at direction (45°, 40°) reduces thepeak gain down to 21.7 dBi and broadens the beam a moderate amount (Figure 6). As
the phasing is changed to steer the beam toward the horizon, the array reaches a limit
due to the patterns of the original patch antennas, and a gain plot with large side lobes
forms. In Figure 7 several beam patterns are shown together to illustrate the steering of
the beam from normal down to 50 degrees in 10-degree steps.

Figure 3: A line plot of the gain in a cross-sectional
cut of the array pattern for the case where all patches are fed in-phase with equal amplitudes. The peak
gain is just over 23 dBi and the 3 dB beamwidth isabout 12 degrees.

Figure 5: The three-dimensional gain pattern for the 8x8array when the patches are phased to direct the main
beam toward (20°, 90°).

Figure 4: The three-dimensional gain pattern for the
8x8 array when all patches are fed in-phase with equal
amplitudes.

Figure 6: The three-dimensional gain pattern for the 8x8array when the patches are phased to direct the main
beam toward (40°, 45°).

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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array

Figure 7: Three-dimensional gain patterns for six gain patterns fromthe 8x8 array for phasing set to direct the beam to (0°,90°) to (50°,90°) in 10-degree increments.

Simulations for return loss were run for each port and were found to have values below-30 dB indicating the patches were properly tuned. Radiation efficiency varied fromabout 78% up to more than 90% over the array, with patches near the edges of thearray generally having higher efficiencies.
To examine the performance of sub-arrays, a few simple cases were simulated to findtypical results for 4x4, 2x2, and 1x2 arrays of elements. All these simulations were
performed with the patches fed with equal phase signals. The patterns for 4x4 and2x2 sub-arrays in a corner of the main array are shown in Figure 8 and Figure 9. Twolocations on the array, one near an edge and one near the center, were simulated as
1x2 sub-arrays and there were only minor differences in their results. A typical pattern
for a 1x2 sub-array defined near the center of the main array is shown in Figure 10.
It is inefficient to simulate all possible phasing combinations to determine the overall
performance of the array. However, a macro program in XFdtd is available that allows afull examination of the gain levels at all angles from the array by sweeping the phasingof the individual elements. The result is a plot showing the cumulative distributionfunction (CDF) of the effective isotropic radiated power (EIRP).

Figure 8: The three-dimensional gain pattern for a
4x4 sub-array of elements in one quadrant of themain array.

Figure 9: The three-dimensional gain pattern for a
2x2 sub-array of elements in one corner of the mainarray.

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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array

Figure 10: The three-dimensional gain pattern for a 1x2 sub-array of elements near
the center of the main array.

The EIRP is an indication of the power an antenna can radiate in a given direction
compared to an isotropic radiator. This plot may be used to determine the fractional
area of the far-zone sphere that has gain above 0 dBi for a given input power level.
Generally speaking, a power level of 23 dBmW is used as the input level for mobile
devices. When the CDF for the entire 8x8 array is generated, it is found that the 23
dBmW level is about 0.225 fractional area (Figure 11), meaning that (1 – 0.225 =0.775) 77.5% of the far-zone sphere can be illuminated with gain above 0 dBi. The 4x4
sub-array (Figure 12) has 64.3% coverage at 23 dBmW input power. Similar plots areshown for the 2x2 sub-array (Figure 13, 50%), and a 1x2 sub-array near the center ofthe main array (Figure 14, 40.2%). Many other sub-arrays beyond those shown here
are possible and may be of value depending on the needs of the system.

Figure 11: The CDF of EIRP plot for the full 8x8 array shows positive gain over 77.5% of the far-zone sphere for an input power of 23
dBmW.

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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array

Figure 12: The CDF of EIRP plot for a 4x4 sub-array located in one
quadrant of the main array showing positive gain over 64.3% of the farzone sphere for an input power of 23 dBmW.

Figure 13: The CDF of EIRP plot for a 2x2 sub-array located in one corner ofthe main array showing positive gain over 50% of the far-zone sphere for an
input power of 23 dBmW.

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Example 7: Beamforming for an 8x8 Planar Phased Patch Antenna Array

Figure 14: The CDF of EIRP plot for a 1x2 sub-array located in near the center of the main array showing positive gain over 40.2% of the far-zone sphere
for an input power of 23 dBmW.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
Summary: This example describes a more developed 28 GHz antenna array capable offorming multiple beams for use in applications such as 5G network base stations.
Introduction
This example describes a more developed 28 GHz antenna array capable of forming
multiple beams for use in applications such as 5G network base stations. The system
consists of three parts: a Rotman lens beamformer with seven input ports and eight
output ports, a series of stripline Wilkinson power dividers to split each Rotman output
into eight equal signals, and an 8x8 patch antenna array. The system will create sevenfocused beams with 3dB beamwidth of approximately 14.5 degrees and greater than 17
dBi Fggain that cover a +/- 30 degree area.
The design process consists of three separate stages: the creation of the Rotman
lens beamformer, the design of the 1 to 8 Wilkinson power divider, and the 8x8 patch
antenna array. The Rotman lens is designed as a microstrip device using Remcom’s
Rotman Lens Designer® (RLD) software. The power divider and the patch array aredesigned in XFdtd®.
This example will describe the creation of each stage of the device and evaluate theperformance of the individual stages and the full device.
Device Design
The Rotman lens beamformer is desired to operate at 28 GHz and to be able to scanseven beams +/- 30 degrees through eight array ports. A 50-ohm microstrip design
is chosen which uses a circular contour shape and an overall width that is just under
5 wavelengths. The sidewalls are curved and include four dummy ports per side forabsorbing any reflected fields. A suitable dielectric is chosen for the substrate with arelative permittivity of 2.94 and a thickness of 0.254 mm. The basic design is shown in
Figure 1 where the RLD software used to create the lens is shown.

Figure 1: The initial Rotman lens design isshown in the RLD software. The seven beam
ports are at the left and the eight array ports
are at the right.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
The beam ports (input) are at the left of the image in grey while the array ports (output)
are at the right in red. The output ports at the ends of the transmission lines are spaced
a half-wavelength apart. The transmission lines are of varying lengths as determined by
the equations of the Rotman lens. The Rotman lens is generally used with one or morebeam ports active to produce a linear phase shift across the array ports due to the timedelay in the signal propagation to reach the output. These devices are often referred toas “true time delay” systems and do not rely on phase shifters to steer beams.
Following tuning of the lens shape, the seven output beams, one per input port, areplotted in Figure 2 to verify the position of the beams and the sidelobe levels. The
beams are present at +/-30, +/-20, +/-10 and 0 degrees of scan angle. A uniform
aperture distribution array ports is intended.

Figure 2: The Array Factor, a measure of the expected radiation pattern produced
by a beam port due to the phase across the array ports, is shown for all sevenbeams of the Rotman lens designed in RLD.

The Rotman lens design from RLD was exported into an SAT format CAD file for use
in XFdtd. Following importation into XFdtd, the resulting geometry of Figure 3 wasdeveloped where all ports are terminated with a 50-ohm load. The return loss for eachbeam port was computed as shown in Figure 4 where acceptable values below -10 dB
are found at 28 GHz. Via a script in XFdtd, the output complex voltages across thetransmission lines connected to the array ports are used to compute the beam pattern
for each input port which are shown in Figure 5. As can be seen, these are quite similar
to the beams from the original RLD design.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G

Figure 3: The Rotman lens design is shown in XFdtd following importation of
a CAD file that was generated by the RLD software program. The lens is donein microstrip on a 0.254 mm substrate with permittivity of 2.94.

Figure 4: The return loss for each beam port of the Rotman lens is plotted overa frequency range around 28 GHz.

Figure 5: This figure shows the Array Factor for
the Rotman lens as simulated by XFdtd. The
complex voltage at the terminal end of eachtransmission line connected to the array ports
was used to produce the Array Factor for each
input port, and it can be seen that these beam
patterns are very similar to those from RLD
shown in Figure 2.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
An analysis of the S-parameter data across the array ports shows that the magnitude
has variations from -11 to -15 dB which are slightly more significant than the desired
uniform distribution (Figure 6). The phase variations across the array ports shownear linear variations, as desired, for each of the seven beams (Figure 7). In Figure
7 the phase is adjusted to put zero phase on array port 4 for comparison. The phasevariation resulting from the Rotman lens between array ports is +/-90, +/-60, +/-30, and0 degrees for the seven beams.

Figure 6: A uniform distribution across the array ports of the Rotman lens was desired in
the design. In the XFdtd simulation, it can be seen that there is some variation in the actual
distribution.

Figure 7: The phase variation across the array ports is plotted for each input
beam port. The expected phase for the designed system should be linear witha slope ranging from +90 degrees between output ports to 0 degrees to -90
degrees depending on which input port is active. Here the phase variations canbe seen to be nearly linear with slopes close to the desired values.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
The stripline Wilkinson power divider is designed using the same dielectric (epsilon= 2.94) as for the Rotman lens. The thickness of the stripline substrate is 0.508 mm
and the traces are 50 ohms. The design splits one input into eight equal and in-phase
outputs in three stages. The output of each port of the Wilkinson is connected via ashort coaxial cable to the input of a patch antenna. The patch antenna array consistsof eight 1x8 subarrays where each subarray is connected to one Wilkinson. The patch
array has a substrate of the same dielectric with 0.254 mm thickness. The patches arespaced a half-wavelength (at 28 GHz) apart and are tuned for best performance with apatch size slightly larger than a quarter wavelength and a feed offset by 0.9 mm fromthe center of the patch. The combined Wilkinson-Array parts are shown in Figures 8
and 9.

Figure 8: Shown in the figure is the three-dimensional CAD
representation of the 8x8 patch antenna array and theeight Wilkinson power dividers that attach to the antennas.
Here the Rotman lens has been replaced by eight input
waveguide ports on the first stage of the Wilkinson power
dividers.

Figure 9: A side view of the Wilkinson power divider is shown which clearly displays the three stagesof dividing the signal.

The combined Wilkinson-Array part was simulated with waveguide ports added toeach of the eight Wilkinson inputs. The resulting return loss plots for each subarrayare shown in Figure 10 and good performance is seen at 28 GHz. The ports are fed
with phase shifts varying from +90 degrees between elements (beam 1) to -90 degrees
between elements (beam 7) in 30 degree increments to produce seven distinct beams.
The beam patterns (plotted as gain rather than Array Factor) are shown in Figure 11. In
Figure 11 there is some slight variation in the peak gain, but the beam locations are welldistributed at the desired angles which are a close match to the original RLD design.
Figures 12 and 13 show the beams in three dimensions along side the geometry
structure. The large white arrows indicate the direction of peak gain.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G

Figure 10: The return loss for each of the eight ports
attached to the Wilkinson power dividers shows good
performance at 28 GHz. There is only slight variation
between the different 1x8 subarrays attached to eachpower divider.

Figure 11: The beam patterns for the combined
Wilkinson and antenna array are plotted as gain
(rather than Array Factor) for the device of Figure 8.
Here the phase relationships that would be producedby the Rotman lens have been replaced by phaseshifts across the input ports attached to the Wilkinson
power dividers. The beams can be seen to be in thecorrect angular position with nearly equal gain.

Figure 12: This figure shows the seven beams as athree-dimensional pattern rather than a line plot like
Figure 11. The large white arrows depict the directionof the peak gain for each beam.

Figure 13: This is an alternate view of the three-dimensional beam patterns of the seven beams generated
by the array.

The final step of the design is to combine the Rotman lens beamformer with the power
divider/antenna array. This structure is shown as a three-dimensional CAD model in
Figures 14, 15, and 16. Here, waveguide ports matched to 50 ohms are used at all
open connections including the seven beam ports and eight dummy ports for reflection
reduction in the lens.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G

Figure 14: Here the complete system of
the Rotman lens input, Wilkinson power
divider stage, and 8x8 patch antenna
array are shown as a three-dimensional
CAD model.

Figure 15: This is a side view of the entire system where
the three stage Wilkinson power divider is more clearlyvisible.

Figure 16: This is a top view of the entire system where
the Rotman lens and array transmission lines are moreclearly visible.

Following simulations with a broad band pulse at each beam port input, a plot of the
return loss versus frequency is shown in Figure 17. The return loss for all ports is goodand near -16 dB with the exception of the central port (beam 4) which has a higher
return loss, perhaps due to the symmetrical location and reflections that are not wellabsorbed in the dummy ports.

Figure 17: Plotted here is the return loss from eachof the seven input ports of the Rotman lens whenconnected to the entire system of power dividers
and antennas. The results are generally good withvalues below -10 dB for all ports except the centerbeam (port 4) which has some mismatch likely due
to reflections not well absorbed by the sidewalls.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
The beam patterns at 28 GHz for the full device are shown in Figure 18 and can be seen
to be properly located in angle with somewhat higher variations in peak gain comparedto the simulations of the array without the Rotman lens. This is due to the less-thanperfect phase and magnitude variations across the Rotman lens array ports that feed
the Wilkinson power divider. The results are consistent with the previous beam patterns
generated in earlier simulations shown in Figures 2, 5, and 11 despite showing morevariation in the beam magnitudes.

Figure 18: The resulting beams from the entire system are plotted as gain patterns
and can be seen to be similar to the other beam pattern plots, with beams spaced
10 degrees apart from -30 to 30 degrees and near equal magnitude.

Three-dimensional views of the beams give a clearer image of the performance ofthe device including the presence of sidelobes and their levels. In Figures 19, 20, 21,
and 22, the first four beams of the device are shown in relation to the device structure.
Beams five, six and seven would be similar to three, two and one, respectively. In Figure
23, all seven beams are displayed in three dimensions to show the full range of coverageof the device. Figure 24 shows the same seven beams from a different viewing direction
located above the device along the Y axis.

Figure 19: Shown in this figure is a three-dimension
view of the beam pattern from input port 1 of the Rotman lens generated by the entire system.

Figure 20: Shown in this figure is a three-dimension view of the beam pattern from input port 2 of
the Rotman lens generated by the entire system.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G

Figure 21: Shown in this figure is a three-dimension
view of the beam pattern from input port 3 of the Rotman lens generated by the entire system.

Figure 22: Shown in this figure is a three-dimension
view of the beam pattern from input port 4 of the Rotman lens generated by the entire system.

Figure 23: This figure shows all seven beams of theentire system as three-dimensional gain patterns.

Figure 24: The seven beams generated by each input
port of the Rotman lens are shown in three dimensions
from a top view.

The visualization of the propagation of conduction currents on the device is a useful toolfor ensuring that all connections between stages are good (no unconnected parts) andthat the propagation speed to the arrays is the same. It is important for the transmission
lines from the Rotman lens to the power dividers be the correct lengths to maintain thephase relationship of the wave front. In Figure 25, the current is seen propagating fromthe center beam port across the lens. By Figure 26 the current has just reached the output of the array ports and is entering the transmission lines. Figure 27 shows the current
fully through the transmission lines and entering the power dividers. In Figure 28, thegeometry display is disabled to better visualize the currents as they propagate throughthe first power divider stage and are still in good phase relation. Finally, in Figure 29 thecurrents have just reached the feeds of the patch antennas and are still in phase witheach other.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G

Figure 25: This figure shows the propagation of conduction currents from beam port 4 into the Rotman lens.

Figure 26: In this figure, the propagation of conduction
currents from beam port 4 of the Rotman lens has justreached the array ports of the lens.

Figure 27: The propagation of the conduction currents
has reached the ends of the array transmission lines
and is entering the Wilkinson power divider elements.
All currents appear to be in phase as they are arriving at
the same location at the same time.

Figure 28: Here the display of the geometry has been
disabled and only the conduction currents on the metal
surfaces are shown. The fields are beginning to split in
the first stage of the Wilkinson power divider.

Figure 29: The currents have finally propagated across the entire structure and havereached the input ports of the antennas.

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Example 8: 28 GHz Beamforming System and Antenna Array for 5G
Conclusion
This example demonstrates one process for generating and analyzing a 28 GHz steerable array for 5G applications. Here the requirements were for beam coverage from -30
to 30 degrees in seven beams which was realized using a Rotman lens beamformer,
eight 1 to 8 Wilkinson power dividers, and an 8x8 array of patch antennas.
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Example 9: 5G Antenna Array Simulation for mmWave Wireless Virtual Reality Headsets
Summary: A 60 GHz antenna array design is simulated in XFdtd to demonstrate
suitability for use on wireless Virtual Reality headsets. The antenna array is comprisedof elements each containing two patches and a parasitic element. The resulting array
produces a fan beam which may be steered by varying the phasing between theelements resulting in broad coverage. The final design is simulated mounted on asection of a virtual reality visor.
Introduction
This example demonstrates an antenna array for 5G 60 GHz applications of wirelesscommunication for wearable devices such as virtual reality headsets. The antenna
design for this example is from the paper [1] by Hong and Choi of Hanyang University.
The array is composed of four elements which each have two patches and a parasitic
element. The parasitic element aids in producing a wider beam in one dimension to
give better coverage. The beams may be steered by varying the phase shift between
elements to provide near hemispherical coverage.
Device Design and Simulation
The proposed array design is shown in Figure 1 where the red material represents
the substrate (Taconic TLY, relative permittivity of 2.2 and loss tangent of 0.0009) andthe green material is Copper. The bottom layer of the antenna is the substrate witha ground plane underneath. This layer is topped by 50-ohm feed lines which thengo through impedance matching before reaching the first patch, then a 70-ohm line
connects to the second patch. Those patch elements are covered by another substratelayer which is topped by parasitic elements that are perpendicular to the 70-ohm lines.
The second layer is visible in the three-dimensional view of the structure in Figure 2.
The elements are spaced a half wavelength apart and are fed by Nodal Waveguideports with variable phase.

Figure 1: Shown is a top view of the antenna arraywith the substrate layers shown in red and themetal feed lines and parasitic elements in green.
The patches are shown as outlines because they
are covered by a second substrate layer.

Figure 2: The antenna array is shown in three
dimensions with the edge of the top substrate layer more
visible over the patches. The four nodal waveguide feed
ports are visible on the top of the array.

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Example 9: 5G Antenna Array Simulation for mmWave Wireless Virtual Reality Headsets
Initially the antenna array is simulated alone for return loss and gain pattern analysis.
For computing S-parameters, each port is simulated individually with the others
terminated in a 50-ohm load. The resulting return loss for each port is shown in Figure
3 and the response from each port is quite similar to the others. The simulations arealso performed without the parasitic elements to demonstrate the impact of including
them on the return loss. As can be seen in Figure 3, the parasitic elements shift theresonance of the antenna higher in frequency and reduce the depth of the S11 null.
When the antenna is simulated with all ports active and in phase, the broadside gain
pattern is computed as shown in Figure 4. Here it can be seen that without the parasitic
element the pattern has deep nulls around +/- 40 degrees which are undesirable. By
including the parasitic elements, the nulls are reduced and the array produces a wide
fan beam. This beam is better shown in three dimensions in Figure 5 where the pattern
can be seen to be broad in the vertical direction and narrow (3-dB beam width of 24
degrees) in the horizontal direction.

Figure 3: The return loss for each element of thearray is quite similar. The addition of the parasitic
element shifts the response higher in frequency
and reduces the null depth.

Figure 4: The gain pattern broadside of the array is fairly
wide with the parasitic elements included. Without theparasitic elements there are undesired nulls present in
the pattern.

Figure 5: The three-dimensional gain pattern
for the equal phase case shows peak gain to
the side with a fan shaped pattern.

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Example 9: 5G Antenna Array Simulation for mmWave Wireless Virtual Reality Headsets
Beam steering is possible with this array by varying the phasing between the elements.
In Figure 6, the array is shown with a beam steered about 30 degrees in the horizontal
by applying a phase shift of 90 degrees between the elements. In Figure 7, sevenpossible beams are shown by varying the phase shift from -90 to 90 in 30-degree steps.

Figure 6: With a phase shift of 90 degrees between theantenna elements the beam shifts about 30 degrees tothe side.

Figure 7: Sweeping the phase shift from -90 to 90 degrees in 30 degree increments generates seven beams
that cover a broad area.

The antenna array is next mounted on a virtual reality headset on a phantom head todemonstrate a possible application, shown in Figure 8. Due to the very large size ofthe head at 60 GHz, a portion of the total problem space, shown in Figure 9, is actuallysimulated for this demonstration. In Figure 10 the primary beam of the array (allelements in phase) is shown compared to the headset and phantom. When a phaseshift of 90 degrees is applied between the elements, the beam shifts to one side by
about 30 degrees as shown in Figure 11. Figure 12 demonstrates both beams at thesame time.
The dissipated power on the surface of the phantom is also of interest in theseapplications, and it can be computed by XFdtd. In Figure 13, the dissipated power fromthe headset-mounted array on the evaluated surface of the phantom model is shown.

Figure 8: The antenna array is shown mounted on a
virtual reality headset. The headset is attached to aphantom head model.

315 S. Allen St., Suite 416 | State College, PA 16801 USA | +1.814.861.1299 phone |Figure 9: Due to the large size of the problem
space, a section of the headset/head model isused for the actual simulations.

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Example 9: 5G Antenna Array Simulation for mmWave Wireless Virtual Reality Headsets

Figure 10: The computed gain pattern for the case of
all elements in phase is shown with a strong gain ofnearly 15 dBi and a fan-shaped beam.

Figure 11: After applying a 90-degree phase shift
between the elements the beam is shifted about 30
degrees to the side.

Figure 12: Both beams with 0- and 90-degree phaseshifts are shown in this image.

Figure 13: The dissipated power on the surface ofthe head phantom from the radiating array is computed by the software and displayed.

Reference:[1] Y. Hong and J. Choi, “60 GHz Array Antenna for mm-Wave 5G Wearable
Applications,” 2018 IEEE International Symposium on Antennas and Propagation and
USNC/URSI National Radio Science Meeting, pp. 1207-1208, 2018.
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