Voltage Control Method with Non-linear Compensation
and DC-Offset Elimination for One-leg T-type Dual
Active Bridge Converter using Multi-Operation Mode
Hayato Higa

Hiroki Watanebe, Keisuke Kusaka and Jun-ichi Itoh

Power Electronics Research Department
MEIDENSHA CORPORATION
Numazu, Shizuoka, Japanhiga-h@mb.meidensha.co.jp

Dept. of Electric Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japanhwatanabe@vos.nagaokaut.ac.jp,
kusaka@vos.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp

Abstract—This paper proposes a voltage control method forone-leg T-type Dual Active Bridge (TDAB) converter in order toachieve a linearization of a current-control characteristic and anelimination of a dc offset in a inductor current at a transient state.
In the conventional Dual Active Bridge (DAB) converter, thecontroller design is complicated due to the nonlinearity of thecurrent-control characteristic and the dc offset of the inductor
current at the phase-shift angle and the operation mode changing.
In the proposed method, the nonlinearity of the current-control
characteristic is compensated by the reverse system model
including the inductance error. In addition, the dc offset of theinductor current is eliminated by updating the phase-shift angle
and changing the operation mode at the peak or bottom of thebase carrier. From experimental results, the ratio between the
output current command and the actual output current agrees
with the ratio between the nominal value and the error value ofthe inductance, i.e. the maximum error of 8.8%. Moreover, thedc offset of the inductor current is prevented when the phaseshift angle and the operation modes are changed.

accordance with the operation point or the operation mode.
Furthermore, the dc offset of the inductor current occurs when
the phase-shift angle and the operation modes are changed.
This dc offset of the inductor current is increased by thedifference between the previous value and the current value ofthe phase-shift angle. Therefore, the change in the amount ofphase-shift angle is limited in order to prevent the magneticflux saturation, i.e. the controller gain has to be designed to be
low.

Keywords—Dual Active Bridge
elimination; Voltage Control Method

Converter;

DC-offset

I.
INTRODUCTION
Recently, a dc micro grid has been proposed for highsystem efficiency [1-4]. This system has numerous isolated dcdc converters which have various demands such as a galvanic
isolation, a wide dc voltage variation, a voltage control, a highefficiency in wide load and a bi-directional operation. A DAB
converter achieves the high efficiency when the input and theoutput dc voltage are close to the rated voltage condition
because zero voltage switching (ZVS) is achieved without
additional components [5-6]. However, the converter efficiency
is drastically decreased during the fluctuation of the input or
the output dc voltage due to the hard switching operation andthe increase of the inductor current [7-8].
In order to improve the converter efficiency against widevoltage variations, multi-level DAB converter using a TDAB
converter [9-10] or a one-leg TDAB converter [11-13] withmulti-mode operation have been proposed. However, thenonlinearity on the phase-shift angle to the output current
occurs in the current-control characteristic of the multi-level
DAB converter. As a result, the controller design iscomplicated because the transient response is varied inThis paper proposes a voltage control method with a nonlinear compensation and a dc-offset elimination in atransformer current for a one-leg TDAB converter. In the oneleg TDAB converter, two operation modes are changed as toimprove the efficiency in wide load and output dc voltage
variation. The proposed voltage control method has following
features; the linearization of the current-control characteristic
including the inductance error regardless of operation modes
and the dc-offset elimination in the transformer current at thetransient state. Consequently, the new contribution of thispaper is the voltage control method with the simple controller
design. This paper is organized as follows; first, the circuit
configuration and the operation principle of the each operation
mode are explained. Second, the nonlinearity compensationand the dc-offset elimination methods are introduced. Finally, a
1.5-kW prototype is demonstrated by the experiment at several
voltage conditions in order to confirm the validity of the
proposed voltage control method.
II.

CIRCUIT CONFIGURATION AND OPERATION PRINCEPLE
OF ONE- LEG TDAB CONVERTER

A. Circuit configulation
Fig. 1 shows the circuit configuration of the one-leg TDAB
converter. The one-leg TDAB converter consists of the DAB
converter, a bidirectional switch Sucp, Sucn, a high frequency
transformer, an external inductance Lex, and two capacitors Cp
and Cn which capacitance are same capacitors.
Fig. 2 shows the operation modes of the one-leg TDAB
converter. A Full bridge (FB) mode and a half bridge (HB)
mode are changed on the input side as shown in Fig. 2 (a) and(b).

Low voltage (LV) side

High voltage (HV) side
Sup

Cp

Vin

iDAB Iout

N1
N2
N1:N2 Swp

Sxp

vLV

iLV

Swn

Sxn

N=

Svp

Lex

l

Sucp Sucn

iHV
vHV

NvLV

Cn

Sun

Sucp,Sucn
Sup,SunCout Vout

Svp,Svn

Lm

Svn

U-phase V-phase

Swp,Swn
Sxp,Sxn

Vin
0
-Vin

vHV

W-phase X-phaseFig. 1. Configuration of one-leg TDAB converter.

1/fsw

H
L
H
L
H
L
H
L



NVout
NvLV

Cp

Sup

Svp

iHV

vHV Vin

Cp

Sup

Svp

iHV

vHV

0
-NVoutiHV_FB() 0
Vin

Sucp Sucn
Cn

Sun

Svn

Sucp Sucn
Cn



Sun

Pattern

Svn





0
I

+
III

II


IV

(a) FB mode (Power flow is HV side to LV side.)Sucp, Sucn: OFF
Sup, Sun,Svp, Svn: Duty ratio 50%

Sucp, Sucn: ON Sup, Sun:OFF
Svp, Svn: Duty ratio 50%(a) Full bridge (FB) mode (b) Half bridge (HB) mode
Fig. 2. Switching patterns of FB and HB mode.

B. Operation analysis with each operation mode
Fig. 3 shows the operation waveforms of the one-leg
TDAB converter. In both HB and FB mode, the output voltage
vHV and NvLV are square waveforms which have two voltage
amplitude, i.e. FB mode; ±Vin, HB mode; ±Vin/2. Thus, thetheoretical equations of the each operation mode are derived in
the same manner with a conventional DAB converter. The
direction of the power transfer is controlled by the phasedifference  between the HV- and LV-side inverters. There are
four patterns in each switching period. As shown in Fig. 3(a)and 3(b), For the sake of brevity, only patterns I and II are
considered here because the patterns III and IV are analyzed in
the same manner. It should be noted that the magnetizing
current is neglected because the total inductance of the leakage
inductance l and the external inductance Lex is much smaller
than the magnetizing inductance Lm. In FB and HB modes, theinductor current during pattern I and pattern II iHV_FB_I, iHV_FB_II,
iHV_HB_I, iHV_HB_II are calculated as
FB mode Pattern


(Vin  NVout )

L



FB mode Pattern


iHV _ FB _ II ( )  iFB _ HV _ I ( ) (Vin  NVout )(   )  L

HB mode Pattern


iHV _ HB _ I ( )  iHV _ HB (0) (Vin / 2  NVout )

L





NVout
NvLV
0
-NVoutiHV_HB() 0

iHVPattern





0
I

II



+
III

IV

(b) HB mode (Power flow is HV side to LV side.)Fig. 3. Operation waveforms of FB mode and HB mode

HB mode Pattern


iHV _ FB _ I ( )  iHV _ FB (0) 

1/fsw

H
L
H
Sup,SunL
H
Svp,Svn
L
Swp,Swn H
Sxp,Sxn L
Vin/2
vHV
0
-Vin/2
Sucp,Sucn

iHV _ HB _ II ( )  iHV _ HB _ I ( ) (Vin / 2  NVout )(   ) 
L

where  is the phase shift angle between the HV- and LV- side
inverters, Vin is the dc voltage of the input side, Vout is the dc
voltage of the load side,  is the angular frequency of theswitching frequency and L is the total inductance of theexternal inductance Lex and the leakage inductance l. In Fig. 3,
the absolute values of iHV_FB_II () and iHV_HB_II () are equal tothat of iHV_FB_I () and iHV_FB_I (), respecitively. Thus, theinitial value of the inductor current iHV_FB_I () and iHV_HB_I ()are calculated by (5) and (6).

iHV _ FB (0)  iHV _ FB _ II ( )




1 

 NVout   (Vin  NVout ) 
L 
2

(5)




 NVout  :Vin  NVout  1 Vin  2

FB mode Vin    1  NV  2 : Vin  NVout
out 








 NVout     1 
 : Vin / 2  NVout Vin / 2  2
HB mode  Vin / 2     1  NV  2 : Vin / 2  NVout
out 





iHV _ HB (0)  iHV _ HB _ II ( )




1 

 NVout   (Vin / 2  NVout ) 
L 
2

(6)

By substuting (5) and (6) into (1) and (3), the inductor
current values of the pattern I with the each operation mode
iHV_FB_Iand iHV_FB_Iare calculated by (7) and (8).

(V  NVout )iHV _ FB _ I ( )  iHV _ FB (0)  in

L

1 


Vin  (Vin  NVout ) 
L 
2



(Vin / 2  NVout )

L

1 


Vin / 2  (Vin / 2  NVout ) 
L 
2



iHV _ HB _ I ( )  iHV _ HB (0) In addition, the root mean square (RMS) value of theinductor current with each operation mode IHV_FB and IHV_HB are
determined by (9) and (10) [11].

i






1

I HV _ FB 

HV _ FB

0












2
2
1 
iHV _ FB _ I ( ) d   iHV _ FB _ II ( ) d     0

2 3
 (Vin  NVout )  2 
3
12NVinVoutL
I HV _ HB 

NVoutVin

L

2

i

1



0

HV _ HB

Ptr _ FB 

NVinVout   
 1   L  



Ptr _ HB 

NVinVout   
 1   
2 L  2( ) dNVinVout

The ZVS range of HB mode is wide when the voltage ratio
Vin / NVout is close to 0.5. Besides, the ZVS range of HB mode
is wide when the voltage ratio Vin / NVout is close to 1.0.
Accordingly, two operation modes are switched in accordancewith the input and the output dc voltage condition and thetransferred power to achieve the high efficiency in wide load.
Then, the transferred powers in each operation mode Ptr_FB and
Ptr_HB are calculated by the phase shift angle between the inputside inverter and the load-side inverter  as (13) and (14) [11].



According to (13) and (14), the transferred power of HB
mode is lower than that of FB mode.
III.

2

2

( ) d 3  2  2 Vin / 2  NVout 



3
2 12
NVoutVin2 In FB mode, the inductor current is minimized at thevoltage condition of Vin = NVout, whereas the inductor current
of HB mode is minimized at the voltage condition of Vin / 2 =NVout. Besides, the ZVS condition is determined by the
inductor current which is varied by the phase-shift angle andthe voltage conditions [7]. In addition, ZVS is achieved in thesame manner of the conventional DAB converter. Thus, theZVS conditions in each operation mode are shown in phaseshift angle as (11) and (12) [11].

VOLTAGE CONTROL METHOD WITH LINEARIZATION AND
DC-OFFSET ELIMINATION

A. Non-linear compensation
Fig. 4 shows control block diagrams of one-leg TDAB
converter. In the power flow from HV side to LV side, i.e. ,the output current of each operation mode are calculated by
dividing (13) and (14) by the output voltage Vout. As a result,the output current of each operation mode iDAB_out are expressedas

iDAB _ out 

Ptr
K NV   2 
 m in    
Vout
L 
 



where Km is coefficient which is decided by the operation
modes. The coefficient Km is 0.5 and 1.0 with HB mode and FB
mode, respectively.
The transfer function of the current-control characteristic in
the one-leg TDAB converter GDAB is established in Fig. 4(a). In(15), the transfer function of the current-control characteristic
in the one-leg TDAB converter GDAB has the nonlinearity
because GDAB changes according to the phase-shift angle. Thisworsens the voltage response when PI controller is applied.

Therefore, the output of the PI controller is necessary to becompensated. In order to compensate the nonlinearity of oneleg TDAB converter, the phase-shift angle  is calculated by(16) to match the output current in two modes.

Plant



NVin
NVin




8 f sw L iDAB _ in   1  1  2
K m NVin 



+


(a) Block diagram for one-leg TDAB converter
iout

The control block diagram in the one-leg TDAB converter
is established in Fig. 4(b). By using (16), the one-leg TDAB
converter’s nonlinearity is eliminated over entire range.
However, the actual inductance Lact might be different from thenominal value L. If the phase-shift angle-to-current transfer
function has the inductance error ratio ker=Lact/L, which isdefined by substituting (15) into (16). The current of controller
side to current-transfer function Gc is expressed asGc iDAB _ out _ FB
iDAB _ in _ FBiDAB _ out _ HBiDAB _ in _ HB

 1/ ker 

According to (17), the transfer function Gc is determined byonly the inductance error ratio ker. As a result, the one-leg
TDAB converter nonlinearity is eliminated by using the reverse
system model regardless of the operation mode and theinductance error.
B. Dc offset in inductor current at operation mode changing
Fig. 5 shows the transient waveform at the conventional
operation mode changing. The dc offset of the inductor current
might occur when the operation mode and the phase-shift angle
are changed. The dc offset of the inductor current is increasedby the difference between the previous value and the current
value of the phase-shift angle. The dc offset of the inductor
current causes the increase of current rating devices andsaturation of magnetic flux in the magnetic component, e.g. theexternal inductance and the high frequency transformer.
Furthermore, the dc offset of the inductor current also occurs at
the phase-shift angle changing [14]. The following switchingsequence eliminates the dc offset of the inductor current at thephase-shift angle and the operation mode changing.
C. Proposed DC-offset elimination method
Fig. 6 shows the transient waveform at the operation modeand phase-shift angle changing. In order to eliminate the dc
offset of the inductor current at the operation mode changing,
The operation-mode-changing timing ch is calculated by (18)when the instantaneous value of the inductor current with FB
mode is equal to that with HB mode.
iHV _ FB _ II ( ch )  iHV _ HB _ II ( ch ) ch 


2





vout

Transfer function of one-leg TDAB GDAB

where iDAB_in is the output current command of the one leg
TDAB converter.

ic 1
iDAB_out
1Km+ - sCout- L
FB mode:Km=1
1
iout R
HB mode:Km=0.5
Eq.(15)

Vout* +

-

vout

+
PI i
c + iDAB_in

Eq.(16)
FB mode:Km=1
HB mode:Km=0.5 1
1
+
+ 
8fswL
|u|
KmNVin√

Reverse system model of one-leg TDAB GDAB-1(b) Control block diagram
Fig. 4. Circuit and control block diagrams of one-leg TDAB converter. In
Fig. 4(a), the current-control characteristic has a nonlinearlity. This leads
the contlloer design is complicated due to changing the transfer function
GDAB. In Fig. .4 (b), the nonliniality compensation is applied by using thereverse system model.
FB mode =3/10

HB mode =3/10U-phasecarrier
Asynchronized to the carrier bottom
Sucp,
Sucn

ON

Operation-mode-changing timing
OFF

vHV 0

vLV 0

1/fsw

DC-offset current

iHV 0
Time

Fig. 5.
Conventional operation mode changing at asynchronous Uphase carrier peak. The dc offset of the inductor current occurs. The dcoffset value is in accordance with the operation-mode-changing timing.

By (18), the operation-mode-changing timing is
synchronized with the peak of U-phase carrier as shown at thetiming B in Fig. 6. In order to realize the phase-shift-angle
commend, the phase-shift carriers for each side are controlled
independently. In the both HV- and LV- carrier, the V-phaseare shifted by leading half of the phase-shift angle commandwhereas the X-phase carrier are shifted by lagging
based on the base carrier On the other hands, the U-phaseare shifted by leading whereas W-phase carrier are shiftedby lagging based on the base carrier. In order to eliminate
DC-offset of inductor current at the phase-shift angle changing,
the phase-shift command of each carrier is updated into twotimes as shown in timing A and C in Fig. 6 because theincrement of the inductor current is reduced by inserting thezero voltage periods. Therefore, the detection of the inductor
current is not necessary to eliminate the dc offset of theinductor current when changing the phase-shift angle and theoperation mode.

D. Control block diagram with proposed method
Fig. 7 shows the control block diagram with the proposed
voltage control method. The proposed voltage control method
has the output current of each operation mode is matched at thechanging timing of the operation mode. The operation-mode –changing flag (Flag_HB) is changed by the relationshipbetween the threshold value and the load current. The thresholdvalue is calculated by the loss calculation of each operation
mode against the voltage condition. In addition, the threshold
HB mode =3/10
Basecarrier/2C

Update timing of 

/2U-phaseV-phaseHV-sidecarrier

/2

/2

IV.

W-phase X-phase

LV-sidecarrierBvHV 0
Operation mode changing timing

B. Stadey state operation
Fig. 8 shows the operation waveform with FB mode at the
rated power and the input voltage of 400 V and the output
voltage of 200 V, i.e. the rated voltage condition. The rated
power operation is confirmed by the experiment.

vLV 0

No DC-offset current
iHV 0

Fig. 9 shows the operation waveform with FB mode and
HB mode at Vin = 400 V and Vout=100 V, i.e. NVout / Vin = 0.5.
In Fig. 9(a) and (b), the amplitude of inverter output voltage ischanged. In addition, the inductor current of HB mode issmaller compared with that of FB mode. Consequently, theinductor current at the light load of 350 W (0.23 p.u.) is
reduced by 39.5%.

1/fsw
Time

Fig. 6. Proposed operation mode and phase-shift changing at
synchronous U-phase carrier peak. The dc offset in the inductor current
does not occur when the operation mode is changed at U-phase-carrier
peak as shown at the timing B. In addition, the phase-shift angle isupdated into two steps of the peak and bottom of the base carrier as shownat the timing A and C.

Fig. 10 shows the output current characteristics of the

DSP

FPGA

dcom=crpk/2
PI0
1|iout|
Ithp
Ithn

Mode
selector

1

MUX
Flag_HBFB
HB

Flag_HB=0:FB mode
Flag_HB=1:HB mode0

1/2

Base carrier

+
-

Sup
Sun

+
+
+
-

U-phase carrier
Update at U-phase carrier peak

Deadtime generator

Km=0.5Detection value
vout, ioutleadinglagging
U and V phaselagging
leading
W and X phase

Generation ofphase-shift carrier

1

Eq.(16) HB

vout

MUX

-

Eq.(16) FB
Km=1.0

Direction ofphase-shift angle

iDAB_in

+

0

Vout*

EXPERIMENTAL RESULTS

A. Laboratory setup
The experimental results are demonstrated in order toevaluate the voltage control method for the one-leg TDAB
converter by using experimental conditions as shown in Table
1. A 1.5-kW prototype (SCH2080KE which is SiC-MOSFET
used at the input side and SCT3030AL which is SiC-MOSFET
used at the load side, Rohm Semiconductor) is operated at theswitching frequency of 80 kHz, the input voltage of 400 V, theload voltage of 200 V to 100 V. In addition, the inductanceerror ratio of ker = 0.8 to 1.2, and the threshold values Ithp, andIthn are 5 A and 4 A. In the operation mode changing, the deadtime between the bi-directional switches Sucp, and Sucn and thearm switches Sup, and Sun is set to 125 ns in order to prevent the
short circuit. It should be noted that the dead-time error in thephase-shift angle occurs in the hard switching operation [14].
In order to compensate the dead-time error, the dead-time-error
compensation method [14] is applied in the experimental set up.

FB mode =/8
Avalue has the hysteresis width Ithp and Ithn to avoid thechattering between FB mode and HB mode. Besides, thephase-shift angle is controlled by the phase-shift carrier which
employs the elimination method of the dc offset of the inductor
current at the load step change in individual operation mode
[14]. Besides, the duty command dcom is set to half of the carrier
peak crpk/2.

Svp
Svn
Swp
Swn
Sxp
Sxn
Sucp
Sucn

Fig. 7.
Control block diagram with changing operation mode. In order to change the operation mode smoothly, the load current of each mode is matchedat the changing timing of the operation mode. In the operation mode changing, the threshold value has the hysteresis width Ithp and Ithn to avoid the chattering
between FB mode and HB mode.

Table I. Experimental conditions.
Inverter output voltage of HV side 1 kV/div
Element

SymbolValue

Rated power

Prated

1.5 kW@Vout=200 V

DC voltage in HV side

Vin

400 V

DC voltage in LV side

Vout

200 V, 100V

Dead time at HV side

Td_HV

125 nsDead time at LV side

Td_LV

100 ns0

External inductanceLex

101 mH

Leakage inductancel23.1 mH

Magnetizing inductanceLm

10.8 mH

Swiching frequency

fsw

80 kHzSampling frequency

fsamp

20 kHz

Propotional gain

Kp

0.2 A/V

Inverter output voltage of LV side 500 V/div
00

Inductor current of HV side 10 A/div
0

Fig. 8.
Operation waveform at rated power of 1.5 kW, Vin = 400 V,
Vout = 200 V
Inverter output voltage of HV side 1 kV/div

0
Integral timeTi

100 ms

Output capacitance

Cout

40 mF

Unit capacitance constant [15]H

267 ms@Vout＝100 V, Pout=750 W

Threshold output current
for switching operation mode

4.5 A ± 0.5 A(Hysteresis width)

Transformer

HV side: Litz wire f0.1*150
LV side: Litz wire f0.1*150*2
N87 ETD 59 (EPCOS) N1:N2=38:19

External Inductor

Litz wire f 0.1*150
N87 ETD 59 (EPCOS)
Gap: 2.5 mm, Turn number: 23

0

Inverter output voltage of LV side 500 V/div

Fig. 11 shows the efficiency characteristics with each
operation mode. In Fig. 11(a), the maximum efficiency of
97.9% is achieved. In the 50% rated output dc voltage of Fig.
11(b), the ZVS is achieved at light load when the HB mode isapplied. Moreover, the converter loss at light load is reduced
by up to 62.3% due to the operation mode changing from FB to
HB mode. In order to achieve the high efficiency in wide load,
the two operation modes are changed at the output power of
500 W.
Fig. 12 shows the loss distribution of the each operation
mode at the voltage condition of Vin = 400 V and Vout = 100 V
in terms of calculated MOSFET losses using the datasheets ofthe HV- and LV-side MOSFETs. In this loss analysis, thelosses of the magnetic component were obtained from themodeling software Gecko MAGNETICS [17]. In this software,
the iron loss is calculated by using the improved-improved
generalized steinmetz equation and improved generalizedsteinmetz equation [18]. The copper loss with the skin andproximity effects is considered in Gecko MAGNETICS [19].
In Fig. 12, the switching loss and the conduction loss with HB

Reduced by 39.5%

IHV=3.06 A

IHV=1.85 A

0

ILV=5.93 A

Inductor current of HV side 10 A/div

ILV=3.70 A

Inductor current of LV side 20 A/div

4ms/div

04ms/div(a)calculation and experimental results. Fig. 10(a) shows the FB
mode results, whereas Fig. 10(b) shows the HB mode results.
In Fig. 10(a) and (b), the ratio between the output current
command and the actual output current agrees with theinductance error ratio, i.e. the maximum error of 8.8%. Thiserror is caused by the phase shift angle error due to theincomplete ZVS [16]. As a result, it is confirmed that thelinearization of the current-control characteristics including theinductance error regardless of the operation mode.

4ms/div

Inductor current of LV side 20 A/div

FB mode at Pout = 350 W (b) HB mode at Pout= 350 W

Fig. 9.
Operation waveforms with each mode at Vin=400 V,
Vout=100 V and 23.3%.

mode are much lower than that with FB mode due to the ZVS
achievement and the reduction of the inductance current at theswitching timing. However, the conduction loss of HV side isincreased because the bidirectional switch in the prototype isused with two MOSFET. In order to improve the prototype
efficiency, the switching device which has a lower onresistance will be selected.
C. Voltage control with operation mode changing
Fig. 13 shows the transient response at the load step change.
This test is conducted with the load step change from 150 W to
750 W. Fig. 13(a) and (c) shows the transient response with the
only FB mode, whereas Fig. 13(b) and (d) shows that with theoperation mode changing. In Fig. 13(a) to (d), the dc offset of
the inductor current is eliminated at the operation modechanging and the phase-angle command shifting. In Fig. 13(b)and (d), the undershoot, overshoot voltage, and the settling
times are 19 V, 24 V, 1.40 ms, and 2.17 ms, respectively. The
maximum error ratio which is compared to only FB mode is
7.0%. Therefore, the response of the proposed voltage control
with changing the operation mode is similar to that with only
FB mode because the nonlinearity in the current-control
characteristic is compensated by the reverse system model
regardless of the operation mode.
V.

CONCLUSION

This paper proposed the voltage control method for the
one-leg TDAB converter in order to achieve the linearizationker=0.8

Efficiency [%]iDAB_out[A]ker=1.24024iDAB_in[A]Maximum efficiency
97.9%@ 1070 W

90

Efficiency
97.3%@ Rated power

85
80

Rated power: 1.5 kW
Operation mode: Only FB mode

75

Maximum error:6.2%
Experimental conditions
Vin=400 V, Vout= 100 V,
Lnom=114 mH, fsw=80 kHz

2

FB mode95

ExperimentCalculation
using Eq.(17)

6

0

100

ker=1.0

8

70

6

8

Changing point

100

Efficiency [%]iDAB_out[A]HB mode95

ker=1.0

Calculation
using Eq.(17)4

ker=1.2

FB modeLoss is reduced
by up to 62.3%.

85

ZVS range
with HB mode80

Operation mode of higher efficiency
75

Experimental conditions
Vin=400 V, Vout= 100 V,
Lnom=114 mH, fsw=80 kHz

HB mode
70

0

200

0
2

3
iDAB_in[A]

4

400

600

800

1000

5

(b) HB modeof the current-control characteristic and the elimination of thedc offset of the inductor current at the transient state. The oneleg TDAB converter with the proposed method contributed thevoltage controller with the simple controller design. As theexperiment results, the converter loss at half of the rated
voltage condition was reduced by up to 62.3%. Besides, theratio between the output current command and the actual
output current agreed with the inductance error ratio, i.e. themaximum error of 8.8%. In addition, the load step response
was confirmed without the dc offset of the inductor current
when the phase-shift angle and the operation mode werechanged. In future work, the converter efficiency will be
improved by the optimizing design of the switching devices
and the magnetic component.
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Inverter output voltage of HV side 1 kV/div
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