A Power Design
Engineer's Guide

MOSFET & GaN FET Application Handbook
A Power Design Engineer’s Guide
Nexperia
Manchester, United Kingdom

MOSFET & GaN FET Application Handbook.
A Power Design Engineer’s Guide
Copyright © Nexperia
www.Nexperia.com
ISBN: 978-0-9934854-7-3
All rights reserved. No part of this publication may be reproduced or distributed inany form or by any means without the prior written permission of the author.
Printed in the United Kingdom

Contributors
Andy Berry
Adam Brown
Dilder Chowdhury
Phil EllisMark Fang
Tony FrielJim Honea
Burkhard Laue
Wayne Lawson
Sami Ould-Ahmed
Kilian Ong
Jim Parkin
Martin Pilaski
Chrisitian Radici
Eleonora Terrugi
Andrew Thomson
Andrei Velcescu
Barry Wynne
Mike ZengPower MOSFETs
Understanding power MOSFET data sheet parameters

1

Designing in MOSFETs for safe and reliable gate-drive operation

2

Using power MOSFETs in parallel

3

Power MOSFET single-shot and repetitive avalanche ruggedness rating

4

Using RC Thermal models

5

Designing RC snubbers

6

Half-bridge MOSFET switching and its impact on EMC

7

Failure signature of electrical overstress on power MOSFETs

8

LFPAK MOSFET thermal design guide9

Maximum continuous currents in Nexperia LFPAK power MOSFETs

10

Thermal resistance (junction to ambient) of LFPAK power MOSFET

11

H-Bridge motor controller design using Nexperia MOSFETs and logic ICs

12Power and small-signal MOSFET frequently asked questions and answers

13

Small-signal MOSFETs
Leakage of small-signal MOSFETs

14

DC-to-DC conversion with small-signal MOSFETs

15

Load switches for mobile and computing applications

16

Level shifting techniques in I²C-bus design

17Power GaN FETs
Understanding Power GaN FET data sheet parameters

18Power GaN technology – Applications, quality, reliability and scalability

19Power GaN technology: the need for efficient power conversion

20

Circuit design and PCB layout recommendations for GaN FET half bridges

21

Probing considerations for fast switching applications

22

Abbreviations
Index
Legal information

Nexperia | A Power Design Engineer's Guide

MOSFET & GaN FET Application Handbook

Introduction
Welcome to the Nexperia MOSFET and GaN FET Application Handbook, the second
edition of our successful design guide. Written by engineers for engineers, thisunique collection of technical materials and application notes provides essential
and up-to-date information for anybody tasked with integrating MOSFETs and
GaN FETs into real-world systems. In the following pages we share expertise andlearnings that Nexperia’s engineering teams have built up over many years ofhelping customers in a variety of sectors take their applications from initial concept,
through prototyping and on into final production.
The knowledge contained in this guide could not be more relevant or timely. Neverbefore has there been such pressure on engineers to create products and systems
that not only deliver high levels of performance and functionality but do so withinincreasingly restrictive size and power constraints. What’s more, the requirement
for cost-effective and efficient power conversion and electrification is only going tointensify as society looks to reduce energy costs, drive down greenhouse gas
emissions and make the most of renewable resources while meeting the increased
demands of a growing population and an explosion in the number of applications
reliant on electricity.
From consumer products to automotive electronics, data centres to industrial
automation, and communication infrastructure to medical equipment, noapplication area is untouched by the need for components that switch, convert andmanage power and engineers who know how to deploy those components tooptimal effect.
At the lower end of the power spectrum, IoT, consumer and other mobile
electronic devices rely on high-efficiency configurations to support the best
possible user experience in tandem with minimizing form factor and maximizingtime between charge or battery replacement. Reliance on electronics for safety,
comfort and infotainment combined with a shift to autonomous, connectedvehicles and hybrid and fully electric powertrains makes power design a criticalfactor in the automotive sector. For data centres significant cost and environmental
benefits can be realized by even small efficiency gains when extrapolated acrossmillions of servers. In modern factory automation systems where output inkilowatts for motor drives is now common, effective use of power is thecornerstone of efficient and accurate motion control. And when it comes tocommunications infrastructure, 5G roll-out is set to generate major demand fortechnologies that deliver high-density, ultra-efficient, ultra-reliable power use.
What is clear is that, irrespective of the application, no longer can power be seen as
an afterthought, something to be addressed only after the design and prototypingof a product or system’s core functionality. What’s more, with efficient andeffective power management at or near the very top of the design agenda, notNexperia | A Power Design Engineer's Guide

only is the role of the power engineer more relevant than ever before but otherengineers are having to broaden their skillset to address power and efficiency
challenges. Which means gaining an understanding of discrete semiconductor
technologies such as power MOSFETs, small-signal MOSFETs and GaN FETs andknowing how best to use these devices to both meet demanding productspecification and performance requirements and deliver the efficiencies expectedby customers and legislators.
For mains- and battery-powered applications, advances in power MOSFET
structures and packaging continue to drive forward system efficiency andperformance. In addition to fast and efficient switching for power supplies, many
power MOSFETs are now designed with particular applications in mind. Suchapplication-specific MOSFETS (ASFETs) may include, for example, optimisedparameters to address soft-start, live insertion, short-circuit resilience, avalancheruggedness and advanced thermal management.
Today’s low-current, small-signal MOSFETs are deployed in applications that range
from DC-DC conversion and load switching to level shifting in bi-directional bus
systems. Among the trends for these devices are high-speed switching andadvanced packaging that supports the performance, power density andminiaturization needs of high-component-density applications such as mobile
battery-powered electronics and wearables.
More recent additions to the portfolio of power discretes include those fabricatedfrom wide bandgap (WBG) materials such as gallium nitride, which have alreadyachieved great success in RF power. Now, thanks to a combination of ultra-low
RDS(on) at high voltages, excellent switching FOM (figure of merit), thermal stability,
high-frequency operation and reducing price points they are becoming important
for a growing number of high-power, high-density, ultra-high-speed switching
applications – including hard-switched topologies where silicon super-junction FETscannot be used. Offering options for automotive, telecommunications, computing
and industrial market sectors, the power GaN technology covered in this Handbook
targets the needs of AC-DC and DC-DC conversion, power factor correction (PFC),
automotive on-board charging and electric drive applications.
It is interesting to note that the patent for the first field effect transistor was filed
by Polish-American physicist Julius E. Lilienfeld in 1926 and the first super-junction
FET was patented back in 1984. Although FET technology has clearly moved on
significantly in the last few decades (not least in terms of switching speeds - someof the latest MOSFETs switch in the time it takes light to move just 3 m), many of
the key issues that engineers must consider remain the same. Understanding theimpact of switching, conduction and avalanche losses on system efficiency, forexample, is critical in most designs, as are techniques for ensuring EMC compliance,
optimised thermal management and reliability. A range of other factors will need
to be taken into account for applications based on emerging GaN FET technology.

MOSFET & GaN FET Application Handbook

Furthermore, as product choice grows, so too does the challenge of narrowing
down the plethora of FET options to those that are most likely to match therequirements of a given use case, comparing those devices on a ‘like-for-like’ basis
and then deciding which will deliver the ‘real-world’ performance demanded by thetarget application. This challenge is not made any easier by a lack of consistency
between suppliers regarding the conditions used for rating key parameters.
This Application Handbook provides useful guidance on all of these topics andmany other issues that the design engineer is likely to encounter when working
with MOSFETs and GaN FETs. By providing insight into their sometimes confusing
and complex behaviour – including information necessary to solve common
problems and avoid potential pitfalls – Nexperia’s belief is that the Handbook will
become a ‘go-to’ reference for anybody tasked with delivering optimised powerand small-signal switching, power conversion and power management.
To supplement the handbook, further product information and the most
up-to-date application notes can be found at www.nexperia.com

Nexperia | A Power Design Engineer's Guide

Contents
Chapter 1
Understanding power MOSFET data sheet parameters
Application Note: AN11158
1

Introduction........................................................................................................... 26

2

Data sheet technical sections............................................................................. 26

2.1

Quick reference data........................................................................................... 27

2.2

Pinning information............................................................................................. 29

2.3

Ordering information.......................................................................................... 29

2.3.1

Marking................................................................................................................... 29

2.4

Limiting values...................................................................................................... 29

2.4.1

Derating curves..................................................................................................... 32

2.4.1.1

Continuous drain current.................................................................................... 32

2.4.2

Power dissipation................................................................................................. 32

2.4.3

Avalanche ruggedness......................................................................................... 34

2.4.4

Safe Operating Area (SOA)................................................................................. 35

2.5

Thermal characteristics........................................................................................ 36

2.6

Electrical characteristics...................................................................................... 36

2.6.1

Static characteristics............................................................................................ 38

2.6.2

Dynamic characteristics....................................................................................... 43

2.6.2.2

Capacitances.......................................................................................................... 46

2.6.2.3

Switching times..................................................................................................... 47

2.6.2.4

Diode characteristics............................................................................................ 47

2.6.2.5

Transfer characteristics....................................................................................... 48

3

Appendices............................................................................................................ 49

3.1

Safe Operating Area (SOA) curves..................................................................... 49

3.1.1

Safe operating area for temperatures above 25 °C....................................... 52

3.1.2

Example using the SOA curve and thermal characteristics........................... 54

3.1.2.1

Calculation steps................................................................................................... 54

3.1.2.2

Derating for higher starting temperatures...................................................... 55

4

References............................................................................................................. 56

5

Tables...................................................................................................................... 57

6

Figures.................................................................................................................... 58

MOSFET & GaN FET Application Handbook

Chapter 2
Designing in MOSFETs for safe and reliable gate-drive operation
Application Note: AN90001
1

Introduction........................................................................................................... 60

2

Data sheet VGS ratings......................................................................................... 61

2.1

VGS threshold voltage - VGS(th)............................................................................. 61

2.2

The maximum VGS ratings - VGS-max.................................................................... 63

3Life testing............................................................................................................. 64

4

The Gate-oxide...................................................................................................... 65

5

Alternative operating profiles............................................................................ 67

5.1

Application assessments..................................................................................... 68

5.1.1

Single MOSFETs.................................................................................................... 68

5.1.2

Half, Full and Three-Phase Bridges.................................................................... 70

5.1.3

High side MOSFETs with Inductive loads.......................................................... 72

5.1.4

Motor Loads.......................................................................................................... 74

6

Application assessment summary..................................................................... 74

7

Creating a mission profile................................................................................... 75

7.1

Example – Full H-bridge motor drive................................................................. 75

8

Summary................................................................................................................ 78

9References............................................................................................................. 79

10

Tables...................................................................................................................... 80

11Figures.................................................................................................................... 81

Chapter 3
Using power MOSFETs in parellel
Application Note: AN11599
1

Introduction........................................................................................................... 84

2

Static (DC) operation............................................................................................ 85

2.1

Worked examples for static operation............................................................. 86

3

MOSFET mounting for good thermal performance and power
sharing.................................................................................................................... 90

4Power sharing in dynamic operation [pulse and Pulse Width
Modulation (PWM) circuits]................................................................................ 92

5

Partially enhanced (linear mode) power sharing............................................ 94

6

Gate drive considerations................................................................................... 95

6.1

Should individual gate drivers be used for each MOSFET in thegroup?..................................................................................................................... 96

7

MOSFET packaging considerations for paralleled groups............................. 96

Nexperia | A Power Design Engineer's Guide

7.1

Bare die (KGD) MOSFETs..................................................................................... 96

7.2

LFPAK MOSFETs.................................................................................................... 96

8

Inductive energy dissipation in paralleled MOSFETs...................................... 97

8.1

Avalanching - low side MOSFET group driving a high sideinductive load........................................................................................................ 97

8.2

Active clamping - high side MOSFET group driving a low sideinductive load........................................................................................................ 98

9

Summary................................................................................................................ 99

10

Tables....................................................................................................................100

11

Figures..................................................................................................................101

Chapter 4Power MOSFET single-shot and repetitive avalance ruggedness rating
Application Note: AN10273
1

Introduction.........................................................................................................104

2

Single-shot and repetitive avalanche definitions..........................................104

3

Understanding power MOSFET single-shot avalanche events...................105

3.1

Single-shot UIS operation..................................................................................105

3.1.1

Single-shot UIS waveforms...............................................................................107

3.2

Single-shot avalanche ruggedness rating.......................................................107

4

Understanding power MOSFET repetitive avalanche events.....................108

4.1

Repetitive UIS operation...................................................................................109

4.1.1

Repetitive UIS waveforms.................................................................................109

4.2

Temperature components................................................................................110

5

Repetitive avalanche ruggedness rating........................................................110

6

Conclusion...........................................................................................................112

7

Examples..............................................................................................................112

7.1

Single-shot avalanche case...............................................................................112

7.1.1

Calculation steps.................................................................................................112

7.2

Repetitive avalanche case.................................................................................112

7.2.1

Calculation steps.................................................................................................112

8

References...........................................................................................................114

9

Figures..................................................................................................................115

MOSFET & GaN FET Application Handbook

Chapter 5
Using RC Thermal models
Application Note: AN11261
1

Introduction.........................................................................................................118

2

Thermal impedance...........................................................................................118

3

Calculating junction temperature rise............................................................119

4

Association between Thermal and Electrical parameters...........................120

5

Foster and Cauer RC thermal models.............................................................120

6

Thermal simulation examples..........................................................................123

6.1

Example 1............................................................................................................123

6.2

Example 2............................................................................................................126

6.3

Example 3............................................................................................................129

7

Discussions...........................................................................................................130

8

Summary..............................................................................................................131

9

References...........................................................................................................131

10

Tables....................................................................................................................132

11

Figures..................................................................................................................133

Chapter 6
Designing RC snubbers
Application Note: AN11160
1

Introduction.........................................................................................................136

2

Test circuit............................................................................................................136

3

Determining CLK and LLK....................................................................................138

4

Designing the snubber - theory.......................................................................139

5

Designing the snubber - in practice.................................................................141

6

Summary..............................................................................................................142

7

Appendix A; determining CLK from Cadd, fRING0 and fRING1.............................142

8

Figures..................................................................................................................144

Chapter 7
Half-bridge MOSFET switching and its impact on EMC
Application Note: AN90011
1

Introduction.........................................................................................................146

2

Double pulse testing..........................................................................................146

2.1

Measurement system........................................................................................148

Nexperia | A Power Design Engineer's Guide

2.2

MOSFET switching simulation..........................................................................149

2.3

MOSFET turn-off waveform description........................................................150

2.4

MOSFET turn-on waveform description.........................................................154

3

Methods to improve switching performance................................................156

3.1

Voltage spikes.....................................................................................................156

3.1.1

Circuit layout.......................................................................................................156

3.1.2

Reducing dI/Dt.....................................................................................................157

3.2

Ringing..................................................................................................................159

3.2.1

Bypass capacitors...............................................................................................159

3.2.2

Adding a snubber................................................................................................160

3.3

Feedback into the gate signal..........................................................................161

3.4

Switching speed control....................................................................................162

3.4.1

Effect of varying RG............................................................................................162

3.4.2

Effect of adding external gate to source capacitance.................................163

3.4.3

Effect of adding gate to drain capacitance....................................................164

4

Impact on EMC and efficiency..........................................................................166

5

Conclusions..........................................................................................................166

6

References...........................................................................................................167

7

Appendix A: Comparison between SPICE simulation and practicalmeasurements for the BUK7S1R0-40H in a double pulse test
circuit....................................................................................................................168

8

Tables....................................................................................................................171

9

Figures..................................................................................................................172

Chapter 8
Failure signature of electrical overstress on power MOSFETs
Application Note: AN11243
1

Introduction.........................................................................................................174

1.1

ESD - Human body model..................................................................................175

1.1.1

EOS method.........................................................................................................175

1.1.2

Fault condition simulated..................................................................................175

1.1.3

Signature..............................................................................................................175

1.2

Unclamped Inductive Switching (UIS) (Avalanche or Ruggedness)............177

1.2.1

EOS method.........................................................................................................177

1.2.2

Fault condition simulated..................................................................................177

1.2.3

Signature..............................................................................................................177

1.3

Linear mode operation......................................................................................179

MOSFET & GaN FET Application Handbook

1.3.1

EOS method.........................................................................................................179

1.3.2

Fault condition simulated..................................................................................180

1.3.3

Signature..............................................................................................................180

1.4

Over-current........................................................................................................181

1.4.1

EOS method.........................................................................................................181

1.4.2

Fault condition simulated..................................................................................182

1.4.3

Signature..............................................................................................................182

2

Appendices..........................................................................................................183

2.1

Human Body Model EOS of BUK9508-55A....................................................183

2.2

Human Body Model EOS of BUK9Y40-55B....................................................185

2.3

Human Body Model EOS of PSMN011-30YL..................................................186

2.4

Human Body Model EOS of PSMN8R5-100PSF.............................................187

2.5

Human Body Model EOS of BUK7Y3R0-40H..................................................188

2.6

Unclamped Inductive Switching EOS of BUK7L06-34ARC...........................189

2.7

Unclamped Inductive Switching EOS of BUK9Y40-55B...............................190

2.8

Unclamped Inductive Switching EOS of PSMN7R0-30YL............................191

2.9

Unclamped Inductive Switching EOS of PSMN8R5-100PSF........................192

2.10

Unclamped Inductive Switching EOS of BUK7Y3R0-40H.............................194

2.11

Linear mode EOS of BUK7L06-34ARC.............................................................195

2.12

Linear mode EOS of BUK9Y40-55B.................................................................197

2.13

Linear mode EOS of PSMN7R0-30YL...............................................................199

2.14

Linear mode EOS of PSMN8R5-100PSF..........................................................201

2.15

Linear mode EOS of BUK7Y3R0-40H...............................................................203

2.16

Over-current EOS of BUK7L06-34ARC............................................................204

2.17

Over-current EOS of PSMN7R0-30YL..............................................................205

3

Tables....................................................................................................................206

4

Figures..................................................................................................................207

Chapter 9
LFPAK MOSFET thermal design guide
Application Note: AN90003
1

Introduction.........................................................................................................212

2

MOSFET power dissipation and the design environment...........................212

2.1

Heat propagation phenomena.........................................................................213

2.1.1

Conduction..........................................................................................................214

2.1.2

Convection...........................................................................................................215

2.1.3

Radiation..............................................................................................................216

Nexperia | A Power Design Engineer's Guide

2.2

Thermal – Electrical analogy.............................................................................217

2.2.1

Thermal resistance.............................................................................................218

2.2.2

Thermal capacitance......................................................................................... 218

2.2.3

Transient and steady state thermal behavior............................................... 219

2.3

MOSFET steady state thermal characteristics.............................................. 220

2.3.1

Thermal resistance junction to mounting base -Rth(j-mb)............................. 220

2.3.2

Thermal resistance junction to ambient - Rth(j-a)........................................... 221

2.3.3

PCB thermal limitation and the 1 Watt rule.................................................. 223

2.3.4

Thermal nomenclature..................................................................................... 224

3

LFPAK56D and LFPAK33................................................................................... 224

3.1

Simple configuration with a single layer....................................................... 224

3.1.1

LFPAK56D........................................................................................................... 224

3.1.2

LFPAK56D only one MOSFET active at a time.............................................. 226

3.1.3

LFPAK33.............................................................................................................. 228

3.2

Usual configuration: 4 layers + vias................................................................ 231

3.2.1

LFPAK56D........................................................................................................... 231

3.2.2

LFPAK56D only one MOSFET active at a time.............................................. 234

3.2.3

LFPAK33.............................................................................................................. 235

3.3

Placement advice for improved dissipation.................................................. 238

3.3.1

LFPAK56D........................................................................................................... 238

3.3.2

LFPAK33.............................................................................................................. 239

3.4

Comparison between two LFPAK56 and one LFPAK56D, then one
LFPAK56D and two LFPAK33........................................................................... 242

3.4.1

Two LFPAK56 to LFPAK56D............................................................................. 242

3.4.2

LFPAK56D to two LFPAK33.............................................................................. 244

3.5

Impact of Rth(j-mb) compared to Rth(mb-a)......................................................... 245

4

LFPAK56 and LFPAK88...................................................................................... 247

4.1

Simple configuration with a single layer....................................................... 247

4.1.1

Set up................................................................................................................... 247

4.1.2

Junction temperature as a function of copper area................................... 248

4.1.3

Maximum allowed power dissipation as a function of copper area......... 249

4.2

Usual configuration: 4 layers + vias................................................................ 251

4.2.1

Set up................................................................................................................... 251

4.2.2

Junction temperature as a function of copper area (4 layers and4.2.3

Maximum allowed power dissipation as a function of copper area

4.3

Simple configuration with a single split layer of copper............................ 255vias)...................................................................................................................... 253(4 layers and vias)............................................................................................... 254

MOSFET & GaN FET Application Handbook

4.3.1

Set up................................................................................................................... 255

4.4

Impact of Rth(j-mb) compared to Rth(mb-amb)..................................................... 257

5Conclusion.......................................................................................................... 260

6

Tables................................................................................................................... 262

7

Figures................................................................................................................. 262

Chapter 10
Maximum continuous currents in Nexperia LFPAK power MOSFETs
Application Note: AN90016
1

Introduction........................................................................................................ 266

2

LFPAK, superior performance......................................................................... 266

3

Key parameters that set out the boundaries................................................ 267

4

MOSFET structure............................................................................................. 268

5

Maximum power and maximum current....................................................... 269

5.1

Maximum power................................................................................................ 269

5.2

Maximum continuous drain current............................................................... 272

5.3

Maximum continuous source current............................................................ 276

6Practical application examples........................................................................ 278

6.1

Output load drive MOSFETs T4, T5, T6 and T7............................................. 279

6.2

Output isolator MOSFETs T2 and T3.............................................................. 281

6.3

Reverse polarity protection MOSFET T1....................................................... 282

7

Tables................................................................................................................... 284

8Figures................................................................................................................. 285

Chapter 11
LFPAK MOSFET thermal resistance - simulation, test and optimization of PCB
layoutApplication Note: AN90019
1

Introduction........................................................................................................ 288

2

Definition of thermal resistance Rth(j-a).......................................................... 288

2.1

Test method....................................................................................................... 288

2.2

Heat transfer in the JEDEC enclosure............................................................ 289

3

MOSFET drain-source current path................................................................ 291

4

PCB layout optimization................................................................................... 292

4.1

PCB trace width calculation............................................................................. 292

4.2

PCB layout simulations..................................................................................... 293

Nexperia | A Power Design Engineer's Guide

5

Conclusion.......................................................................................................... 297

6

Tables................................................................................................................... 299

7Figures................................................................................................................. 299

Chapter 12
H-bridge motor controller design using Nexperia discrete semiconductors andlogic ICs
Application Note: TN90002
1

Introduction........................................................................................................ 302

2

Block diagram and system functionality........................................................ 303

2.1

Subsystems overview........................................................................................ 303

3

Subsystem descriptions.................................................................................... 304

3.1

Power supply...................................................................................................... 304

3.2

Clock and duty cycle generator....................................................................... 307

3.3

Pulse Width Modulation................................................................................... 310

3.4

Drive circuit......................................................................................................... 313

3.5

H-bridge.............................................................................................................. 316

3.6

Overcurrent detection...................................................................................... 318

4

PCB top view.......................................................................................................321

5

Schematics...........................................................................................................321

5.1

Power supply.......................................................................................................322

5.2

Clock.....................................................................................................................322

5.3

Duty cycle.............................................................................................................323

5.4

PWM......................................................................................................................323

5.5

Overcurrent detection.......................................................................................324

5.6

H-bridge...............................................................................................................324

5.7

Driver circuit........................................................................................................325

6

Bill of materials (Nexperia parts).....................................................................326

7

Bill of materials...................................................................................................328

8

Tables....................................................................................................................334

9

Figures..................................................................................................................335

MOSFET & GaN FET Application Handbook

Chapter 13
Power and small-signal MOSFET frequently asked questions and answers
Technical Note: TN00008
1

Introduction.........................................................................................................338

2

Gate.......................................................................................................................338

3

Thermals...............................................................................................................340

4

MOSFET body diode...........................................................................................347

5

Safe operating area and linear mode operation...........................................349

6

Avalanche ruggedness and Unclamped Inductive Switching (UIS)............354

7

Capacitive dV/dt issues......................................................................................362

8

Package and mounting......................................................................................371

9

SPICE models.......................................................................................................373

10

MOSFET silicon technology..............................................................................374

11

Supply and availability........................................................................................376

12

EMC and ESD.......................................................................................................376

13

Leakage, breakdown and MOSFET characteristics......................................380

14

MOSFET reliability..............................................................................................394

15

References...........................................................................................................396

16

Tables....................................................................................................................397

17

Figures..................................................................................................................398

Chapter 14
Leakage of small-signal MOSFETs
Application Note: AN90009
1

Introduction.........................................................................................................402

2

Leakage currents................................................................................................402

2.1

Drain-source leakage.........................................................................................402

2.2

Gate to Source leakage......................................................................................404

3

Applications.........................................................................................................406

3.1

Load switch..........................................................................................................406

4

Selecting a low leakage MOSFET.....................................................................408

5

Nexperia low leakge MOSFET portfolio.........................................................408

6

References...........................................................................................................409

7

Tables....................................................................................................................410

8

Figures..................................................................................................................411

Nexperia | A Power Design Engineer's Guide

Chapter 15
DC-to-DC conversion with small-signal MOSFETs
Application Note: AN11119
1

Introduction.........................................................................................................414

2

DC-to-DC conversion methods.........................................................................414

2.1

DC-to-DC down-converter.................................................................................414

2.2

DC-to-DC up-converter......................................................................................420

2.2.1

DC-to-DC up- and down-converter .................................................................422

3

Medium power DC-to-DC down-converter using small-signal
MOSFETs..............................................................................................................423

3.1

DC-to-DC down-converter application board................................................423

3.2

Nexperia small-signal MOSFETs suitable for DC-to-DC conversion...........426

3.3

Dimensioning aspects for the inductor and output capacitor....................426

3.4

MOSFET losses calculation................................................................................428

4

Figures..................................................................................................................432

Chapter 16
Load switches for mobile and computing
Application Note: AN90017
1

Types of load switches.......................................................................................434

1.1

P-channel MOSFET load switch........................................................................434

1.2

N-channel MOSFET load switches...................................................................435

1.3

Reverse current protected load switches......................................................436

1.4

Reverse polarity protection and supply or-ing..............................................438

2

Important parameters.......................................................................................439

3

Figures..................................................................................................................441

Chapter 17
Level shifting techniques in I2C-bus design
Application Note: AN10441
1

Introduction.........................................................................................................444

2

Bidirectional level shifter for Fast-mode and Standard-mode
I2C-bus systems...................................................................................................444

2.1

Connecting devices with different logic levels.............................................444

2.1.1

Operation of the level shifter...........................................................................445

3

Figures..................................................................................................................447

MOSFET & GaN FET Application Handbook

Chapter 18
Understanding Power GaN FET data sheet parameters
Application Note: AN90005
1

Introduction.........................................................................................................450

2

Nomenclature.....................................................................................................450

3

Pinning information...........................................................................................450

4Two-chip, integrated, normally-off power switch.........................................451

5GaN FET limiting values.....................................................................................452

5.1

Drain-source voltage, VDS..................................................................................453

5.2

Transient drain-source voltage, VTDS...............................................................454

5.3

Gate source voltage, VGS and total power dissipation, Ptot.........................455

5.4

Continuous and pulsed currents, ID, IDM, IS and ISM........................................456

6

GaN FET static characteristics...........................................................................457

6.1

Gate-source threshold voltage, VGS(th)............................................................457

6.2

Drain-source on-state resistance, RDS(on)........................................................4577

GaN FET dynamic characteristics.....................................................................459

7.1

Gate charge QG(tot), QGS and QGD......................................................................459

7.2

Capacitances Ciss, Coss and Crss..........................................................................459

7.3

Effective output capacitance (energy related), Co(er)...................................460

7.4

Effective output capacitance (time related), Co(tr)........................................461

7.5

Output charge and stored energy, Qoss and Eoss...........................................463

7.6

GaN FET switching time characteristics..........................................................463

7.7

Source-drain voltage, VSD..................................................................................464

8

Qr for GaN FET switches....................................................................................465

9

Switching-node snubber...................................................................................466

10

Tables....................................................................................................................467

11

Figures..................................................................................................................468

Chapter 19
An insight into Nexperia Power GaN technology – Applications, quality,
reliability and scalability
Application Note: TN90004
1

Introduction.........................................................................................................470

2

Product robustness............................................................................................471

3

Quality / qualification of power GaN technology including failuremodes.................................................................................................................. 471

3.1

HTRB.................................................................................................................... 472

Nexperia | A Power Design Engineer's Guide

3.2

HTOL.................................................................................................................... 472

4Conclusion.......................................................................................................... 474

5References.......................................................................................................... 475

6

Figures................................................................................................................. 476

Chapter 20
Power GaN technology: the need for efficient power conversion
Application Note: AN90021
1

Introduction........................................................................................................ 478

2Power GaN FET switches.................................................................................. 479

3

Scalability and growth...................................................................................... 483

4Applications and performance........................................................................ 483

5

Quality and reliability........................................................................................ 485

6References and further reading...................................................................... 486

7

Tables................................................................................................................... 487

8Figures................................................................................................................. 488

Chapter 21
Circuit design and PCB layout recommendations for GaN FET half bridges
Application Note: AN90006
1

Introduction - Diode-free bridges................................................................... 490

2Circuit-design recommendations.................................................................... 492

2.1

Solutions to suppress oscillation.................................................................... 493

2.2

Gate drive and bootstrap supply.................................................................... 494

2.3

Summary of circuit-design recommendations.............................................. 494

2.4

Required and recommended external components................................... 495

2.5

To verify GaN FET stable operation............................................................... 495

3

PCB layout........................................................................................................... 496

3.1

Pin assignment................................................................................................... 496

3.2

Power loop.......................................................................................................... 496

4Heat Sink connection........................................................................................ 500

4.1

Gate-drive loop.................................................................................................. 501

4.2

Summary of PCB layout recommendations.................................................. 502

5

Tables................................................................................................................... 503

6

Figures................................................................................................................. 504

MOSFET & GaN FET Application Handbook

Chapter 22
Probing considerations for fast switching applications
Application Note: AN90004
1

Introduction........................................................................................................ 506

2

Oscilloscope probe ground spring................................................................. 506

3

Oscilloscope and probes.................................................................................. 508

4

Probe positioner................................................................................................ 509

5Figures................................................................................................................. 510

Abbreviations................................................................................................................... 512

Index.............................................................................................................................. 516

Legal information............................................................................................................ 536

Understanding power MOSFET data sheet parameters

1

24
Nexperia | A Power Design Engineer's Guide

Chapter 1
Understanding power
MOSFET data sheet
parameters
Application Note: AN11158

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

25

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

1 Introduction
This chapter describes the content of power MOSFET data sheet parameters. The
goal is to help an engineer decide what device is most suitable for a particularapplication.
It is important to pay attention to the conditions for which the parameters arelisted, as they can vary between suppliers. These conditions can affect the values ofthe parameters making it difficult to choose between different suppliers.
Throughout this document, the data sheet for BUK7Y3R5-40H is used as an
example. BUK7Y3R5-40H is an automotive-qualified part in a SOT669 (LFPAK56)
package, with a voltage rating of 40 V.
The layout of this data sheet is representative of the general arrangement of
Nexperia power MOSFET data sheets.
Nexperia Power MOSFETs are designed with particular applications in mind. For
example, switching charge is minimized where switching losses dominate, whereas
on-resistance is minimized where conductive losses dominate.
• General description, describing the technology used, the package and relevant
qualifications, e.g. AEC-Q101• Features and benefits, listing important features of the MOSFET and the benefit
they offer.
• Applications, listing the applications for which the MOSFET is particularity
suited.
The product overview is followed by these technical sections:• Quick reference data
• Pinning information• Ordering information• Limiting values• Thermal characteristics• Electrical characteristics• Package outline

2 Data sheet technical sections
Nexperia power MOSFET data sheets begin with an overview of the device, giving
the designer the key information regarding device suitability. The overview consistsof:• General description, describing the technology used, the package and relevant
qualifications, e.g. AEC-Q101

26

• Features and benefits, listing important features of the MOSFET and the benefit
they offer.
• Applications, listing the applications for which the MOSFET is particularity suited.
The product overview is followed by these technical sections:• Quick reference data
• Pinning information• Ordering information• Limiting values• Thermal characteristics• Electrical characteristics• Package outline

2.1 Quick reference data
The quick reference data table contains more detailed information and the key
parameters for the intended application. An example of a quick reference datatable is shown in Table 1 “Quick reference data”.

Table 1: Quick reference data
Symbol Parameter

Conditions

Min

Typ

Max Unit

VDS

drain-source voltage

25 °C ≤ Tj ≤ 175 °C

-

-

40

V

ID

drain current

VGS = 10 V; Tmb = 25 °C;[1]Figure 1

-

-

120

A

Ptot

total power dissipation

Tmb = 25 °C;
Figure 2

-

-

115

W

VGS = 10 V; ID = 20 A; Tj= 25 °C

2

2.9

3.5

mΩ

ID = 20 A; VDS = 44 V;
VGS = 10 V

-

6

15

nC

IS = 25 A; dIS/dt = -100
A/μs;VGS = 0 V;
VDS = 20 V

-

16

-

nC

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MOSFET & GaN FET Application Handbook

Static characteristics
drain-source on-state

RDS(on) resistance

Dynamic characteristics
QGD

gate-drain charge

Source-drain diodeQr

recovered charge[1] 120 A continuous current has been successfully demonstrated during application tests. 		
Practically the current will be limited by PCB, thermal design and operating temperature.

27

Understanding power MOSFET data sheet parameters

128

Nexperia | A Power Design Engineer's Guide

The general format for describing a parameter is to provide the official symbol andthen the correct parameter name. Any relevant conditions and information arelisted after the parameter names. The values and units of the values are entered in
the remaining columns. Generally, measurement methods are as described in
IEC 60747-8.
The quick reference data parameters are described in more detail in thecharacteristics section of the data sheet. The following list is an introduction tosome of the key symbols together with a description of the parameter each
represents:• VDS - the maximum voltage between drain and source that the device isguaranteed to block in the off state. This section of the data sheet deals with themost commonly used temperature range, as opposed to the full temperature
range of the device.
• ID - the maximum continuous current the device can carry with the mounting
base held continuously at 25 °C with the device fully on. In the example provided
in Table 1, ID requires a VGS of 10 V.
• Ptot - the maximum continuous power the device can dissipate with themounting base held continuously at 25 °C.
• RDS(on) (drain-source on state resistance) - the typical and maximum resistance ofthe device in the on-state under the conditions described. RDS(on) varies greatly
with both Tj and the gate-source voltage (VGS). Graphs are provided in the data
sheet to assist in determining RDS(on) under various conditions.
• QGD (gate-drain charge) - an important switching parameter that relates toswitching loss, along with QGS and QG(tot). QGD is inversely proportional to RDS(on),therefore choosing an appropriate balance between RDS(on) and QGD is critical for
optimal circuit performance.
• Qr (recovered charge) – the total amount of charge recovered from the antiparallel diode when it is switched from its conducting state to its reverse biased
state under controlled conditions. Qr is an important factor involved in voltage
spiking when interacting with external inductances and an importantconsideration when investigating EMC and efficiency. Generally the higher the Qr
value the larger the voltage and current spikes at switch off leading to longer
damping times. Nexperia includes stored charge Qs as well as output charge Qoss
in its stated value for Qr such that Qr = Qs + Qoss.

2.2 Pinning information
This section describes the internal connections and general layout of the device.
Note that the symbol is for an enhancement mode n-channel MOSFET with thesource and body tied together, and a parallel diode between the source and drain.
The parallel diode is known as the body diode and is inherent in power MOSFETs.
N-channel power MOSFETs have the body diode between drain and source, as
shown in Table 2.

Table 2: Pinning Information
Pin Symbol

Description

1S

source2Ssource3Ssource

4

G

gate

mb D

mounting base;connected to drain

Simplified outline

Graphic symbol

D

G

LFPAK56; PowerSO8 (SOT669)

mbb07

S

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

2.3 Ordering information
The ordering section provides information on how to order the device. The package
version and decription are given as well as any commonly used package name.

2.3.1 Marking
Depending on the device package, the data sheet may include a marking section.
This provides the marking code which is printed onto the device during manufacter,(else the device name will be printed).

2.4 Limiting values
The limiting values table provides the range of operating conditions allowed for the
MOSFET. The conditions are defined in accordance with the absolute maximum
rating system (IEC 60134). Operation outside of these conditions is not guaranteed,so it is recommended that these values are not exceeded. Doing so runs the risk of
immediate device failure or reduced lifetime of the MOSFET. The avalancheruggedness conditions, when given, describe the limited conditions for which the
VDS rating can be exceeded.

29

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

To calculate how the limiting values change with temperature, they are read
together with the derating curves provided.
The limiting values table for the BUK7Y3R5-40H is given as an example of astandard limiting values table, in Table 3.

Table 3: Limiting values
In accordance with the Absolute Maximum Rating System (IEC 60134).

SymbolParameter

Conditions

Min

Max

Unit

VDS

drain-source voltage

25 °C ≤ Tj ≤ 175 °C

-

40

V

VGS

gate-source voltage

-10

20

V

Ptot

total power dissipation Tmb = 25 °C;

-

115

W

ID

drain current

VGS = 10 V; Tmb = 25 °C;
Figure 1

[1]

-

120

A

VGS = 10 V; Tmb = 100 °C;
Figure 2

[1]

-

93

A

-

526

A

pulsed; tp ≤ 10 µs; Tmb =
25 °C

IDM

peak drain current

Tstg

storage temperature

-55

175

°C

Tj

junction temperature

-55

175

°C

-

120

A

-

526

A

ID = 120 A; Vsup ≤ 40 V;
RGS = 50 Ω; VGS = 10 V;[2] [3] Tj(init) = 25 °C; unclamped;
Figure 3

45

mJ

Source-drain diodeIS

source current

Tmb = 25 °C

ISM

peak source current

pulsed; tp ≤ 10 µs; Tmb =
25 °C

[1]

Avalanche ruggedness

EDS(AL)S

non-repetitive
drain- source
avalanche energy[1] 120 A continuous current has been successfully demonstrated during application tests. 		
Practically the current will be limited by PCB, thermal design and operating temperature.
[2] Single-pulse avalanche rating limited by maximum junction temperature of 175 °C.
[3] Refer to application note AN10273 for further information.

• VDS (drain-source voltage) - the maximum voltage the device is guaranteed toblock between the drain and source terminals in the off-state for the specified
temperature range. For the BUK7Y3R5-40H, the temperature range is from
+25 °C to +175 °C. For operation below 25 °C, the VDS rating reduces due to thepositive temperature coefficient of avalanche breakdown. This is covered in

30Section 2.4.1 of this document.
• VGS (gate-source voltage) - the maximum voltage the device is specified to block
between the gate and source terminals. Some Nexperia data sheets specify
different values for DC and pulsed VGS. In these cases the DC value is a constantgate voltage over the lifetime of the device at the maximum Tj, whilst thehigher-value pulsed-rating is for a shorter, specified accumulated pulse duration
at the maximum specified Tj.
Gate-oxide lifetime (refer to AN90001 (chapter 2 of this book) - Designing in
MOSFETs for safe and reliable gate-drive operation) reduces with increasing
temperature and/or increasing gate voltage. This means that VGS lifetimes or
ratings quoted for lower junction temperatures are significantly greater than if
specified at higher temperatures. This can be important when comparing data
sheet values from different manufacturers.
• VDGR (drain-gate voltage) is typically the same value as the VDS rating. Thisparameter appears in the data sheets of older devices but is not quoted fornewer devices such as in the BUK7Y3R5-40H data sheet.
• ID (drain current) - the maximum continuous current the device is allowed tocarry under the conditions described. This value can be related to either package
construction, or the maximum current that would result in the maximum Tj. Assuch it depends on an assumed mounting base temperature (Tmb), the thermal
resistance (Rth) of the device, and its RDS(on) at maximum Tj. Note that some
suppliers quote the "theoretical" silicon limit, while indicating the package
limited value in the characteristic curves.
• IDM (peak drain current) is the maximum drain current the device is allowed tocarry for a pulse of 10 μs or less.
• Ptot (total power dissipation) is the maximum allowed continuous power
dissipation for a device with a mounting base at 25 °C. The power dissipation is
calculated as that which would take the device to the maximum allowed junction
temperature while keeping the mounting base at 25 °C. In reality, it is difficult tokeep the mounting base at this temperature while dissipating the 105 W that isthe calculated power dissipation for the BUK7Y3R5-40H. In other words, Ptot
indicates how good the thermal conductivity of the device is, and its maximum
allowed junction temperature.
Note that some other semiconductor vendors quote performance when
mounted on a copper PCB usually 1 inch square. In practice, this information israther meaningless as the semiconductor vendor has no control over how thedevice is cooled. See AN10874 - (chapter 9 of this book) - LFPAK MOSFET thermal
design guide. AN10874 describes different techniques that can be used during
the design phase to ensure that the PCB layout provides optimum thermal
performance.
• Tstg (storage temperature) is the temperature range in which the device can be
stored without affecting its reliability. Long term storage should be in an inert
atmosphere to prevent device degradation, for example, by tarnishing of themetal leads.
• Tj (junction temperature) is the operational temperature range of the device.
Typically, Tj is the same as the storage temperature. Outside of this range, device

1
Understanding power MOSFET data sheet parameters

MOSFET & GaN FET Application Handbook31

Understanding power MOSFET data sheet parameters

1

Nexperia | A Power Design Engineer's Guide

•

•
•

•

parameters are outside the range of the data sheet and device lifetime is 		
reduced.								IS (source current) - the maximum continuous current of the MOSFET body
diode, which is briefly discussed in Section 2.2. The same considerations apply as
for ID.
ISM (peak source current) - the maximum current pulse that the MOSFET body
diode is diode is guaranteed to carry. The same considerations apply as for IDM.
EDS(AL)S (non-repetitive drain-source avalanche energy) - the maximum allowed
single overvoltage energy pulse under the conditions specified. For this example,
the conditions are the maximum continuous drain current allowed for amounting base temperature of 25 °C. The avalanche energy allowed is theenergy pulse that would raise the device temperature from 25 °C to its maximum
allowed Tj, while the mounting base temperature is held at 25 °C.
The avalanche energy is specified for the maximum continuous drain current.
Some vendors specify the avalanche energy for a different current and higher
inductive load, which can increase the apparent avalanche energy for an inferior
performance. An example is given with the derating curve as described in
Section 2.4.3 of this document.
EDS(AL)R (repetitive drain-source avalanche energy) - the maximum amount ofenergy allowed in each avalanche event when more than one avalanche event
occurs. The thermal constraints imposed for repeated avalanche operation is
given by curve 3 of Figure 3 in Section 2.4.3 of this document. There are also thestandard thermal requirements in addition to the energy requirements for
repetitive avalanche events. These requirements are assessed with the thermal
characteristic curves as described in Section 2.5. Avalanche performance iscovered in detail in application note AN10273 (Chapter 4 of this book) - PowerMOSFET single-shot and repetitive avalanche ruggedness rating.
This parameter is only listed on Nexperia data sheets where the repetitive
avalanche capability has been assessed. It is not shown in Nexperia data sheets
where it has not been assessed, for example non-automotive MOSFETs.

2.4.1 Derating curves
The derating curves are provided immediately after the tabulated limiting value
data, and help the designer calculate how the limits change with temperature.

2.4.1.1 Continuous drain current
The following procedure serves as an example to calculate the maximum
continuous drain current for the BUK7Y3R5-40H. Assume an application with amounting base temperature Tmb of 75 °C.
Refer to the graph depicted in Figure 1 which depicts the continuous drain current
as a function of mounting base temperature.
Figure 1 shows that for a Tmb of 75 °C, the maximum continuous drain current has

32reduced from 120 A, listed at 25 °C, to 105 A, see Equation (1) and Equation (2)below.
The maximum current at any Tmb, is the current that increases Tj to the maximum
allowed temperature (175 °C). P = I2 x RDS(on) represents the power dissipation at Tj,
where the RDS(on) used is the maximum value for the maximum Tj. Therefore, theallowed current is proportional to the square root of the allowed power
dissipation.
The power dissipation allowed for a given Tmb is proportional to the allowed
temperature increase. This means that the derating curve shown, is based on thefollowing equations:

T j T mb							I D 2(T mb )T j 25°C[1]		

Tj Tmb							I D (T mb ) = I D (25°C)
T j 25°C[2]

At the maximum allowed junction temperature of 175 °C, this current has
decreased to zero

ID

150

((A)
A)
125

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

(1)

100

75

50

25

0

VGS ≥ 10 V
0

25

50

75

100

125

150 175
Tmb (°C)

200
aaa-026228(1) 120 A continuous current has been successfully demonstrated during application tests. Practicallythe current will be limited by PCB, thermal design and operating temperature.
Figure 1 | Continuous drain current as a function of mounting base temperature

33

Understanding power MOSFET data sheet parameters

1

Nexperia | A Power Design Engineer's Guide

2.4.2 Power dissipation
Power dissipation varies with different temperatures. However, in this case, thepower dissipation curve is normalized. The allowed power is presented as apercentage of the allowed power dissipation at 25 °C, as opposed to an absolutevalue.

Example:By observing the curve in Figure 2, the allowed power dissipation for a Tmb of 75 °C
is approximately 66 % of that allowed at 25 °C. The graphic data in Figure 2, shows
the maximum continuous power dissipation (Ptot) at 25 °C is 105 W.
This means that the maximum power dissipation allowed at 75 °C, is 66 % of 105 W
which is 70 W.
Equation (3) is the equation to calculate power dissipation:
Tj _ Tmb_________
Ptot (Tmb) - Ptot (25ºC) x			
Tj _ 25ºC[3]

Where Tj = Tj(max), usually 175 °C.
120
Pder
(%)
80

40

0

0

50

100

150

Tmb (°C)

200
03na19Figure 2 | Normalized total power dissipation as a function of mounting base temperature

The curves provided in Figure 1 and Figure 2 are read in conjunction with thelimiting values tables. The information extracted, assists in calculating the
maximum current allowed and the power dissipation with respect to temperature.

34

2.4.3 Avalanche ruggedness
Avalanche ruggedness is covered in detail in AN10273 (chapter 4 of this book) Power MOSFET single-shot and repetitive avalanche ruggedness rating.
103

maximum avalanche pulse
120 A for 14.5 µs

IAL
(A)

avalanche pulse
40 A for 150 µs

102

(1)

10
(2)

(3)

1

10-1
10-3

10-2

10-1

1
tAL (ms)10
aaa-026229

Figure 3 | Avalanche current as a function of avalanche period

A simple example for the BUK7Y3R5-40H, using the information in AN10273, isextracted from the limiting values Table 3:

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

With ID = 120 A, Vsup ≤ 40 V, RGS = 50 Ω, VGS = 10 V and Tj(init) = 25 °C unclamped, themaximum EDS(AL)S is 45 mJ.
An avalanche event has a triangular pulse shape, so the average power is calculated
as (0.5 × VDS × IDS).
AN10273 states that the assumed breakdown voltage is 130 % of the rated
voltage, in the case of the BUK7Y3R5-40H this is 52 V (40 V x 1.3).
Figure 3 shows a maximum current of 120 A at 25 °C (the limiting values Table 3
confirms this value). The time for the maximum avalanche energy can be read from
Figure 3 as 14.5 μs.
This means that the maximum avalanche energy allowed is:
0.5 × (40 V × 1.3) × 120 A × 14.5 μs = 45.24 mJ.
However the limit value quoted in Table 5 of the data sheet is rounded to 45 mJ.
If a competitor quotes avalanche energy at 40 A, the graph shows that theavalanche time for the BUK7Y3R5-40H has increased to 150 μs. The avalancheenergy is now
0.5 × (40 V × 1.3) × 40 A × 150 μs = 156 mJ, which is much higher than data sheet

35

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

limiting value. Ruggedness events lie outside the Safe Operating Area (SOA).

2.4.4 Safe Operating Area (SOA)The Safe Operating Area (SOA) curves are some of the most important on the data
sheet.
103
ID(A)

Limit RDSon= VDS / ID

102

tp = 10 us
100 us10
DC1

Spirito effect (uneven die heating) occurs when a longer
pulse causes a larger current to develop and is evident
by a gradient change in the SOA curve

1 ms
10 ms
100 ms

10-1
10-1

1

10

VDS (V)102

aaa-026230

Figure 4 | Safe operating area; continuous and peak drain currents as a function of drain-source
voltage

The SOA curves show the voltage allowed, the current and time envelope ofoperation for the MOSFET. These values are for an initial Tmb of 25 °C and a single
current pulse. This is a complex subject which is further discussed in the appendix(Section 3.1).

2.5 Thermal characteristics
This section describes the thermal impedance as a function of pulse duration fordifferent duty cycles. This information is required to determine the temperaturethat the silicon reaches under particular operating conditions, and whether it iswithin the guaranteed operation envelope.
The thermal characteristics are shown in Table 4. The thermal impedance changes
with pulse length because the MOSFET is made from different materials. For
shorter durations, the thermal capacity is more important, while for longer pulses,
the thermal resistance is more important.
The thermal characteristics are used to check whether particular power loading
pulses above the DC limit would take Tj above its safe maximum limit. Repetitive
avalanche pulses must be considered in addition to the constraints specific toavalanche and repetitive avalanche events.

36

Table 4: Thermal characteristics
SymbolParameter

Conditions

Rth(j-mb)

thermal resistance fromjunction to mounting
Figure 5
base

Min

Typ

Max

Unit

-

1.18

1.3

K/W

P

δ=

10
Zth(j-mb)(K/W)
1

δ = 0.5
0.2

10-1

0.1
0.05
0.02

single shot

10-2

T

t

tp
10-3
10-6

tp

T
10-5

10-4

10-3

10-2

10-1

tp (s)

1

aaa-026231

Figure 5 | Transient thermal impedance from junction to mounting base as a function of pulse

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

duration.

Thermal resistance (Rth) and thermal impedance (Zth) are related because thethermal resistance is the steady-state measure of how the device blocks heat flow.
Thermal impedance is how the device responds to transient thermal events. It
involves different thermal capacities of parts of the device and the thermal
resistances between these parts. Under DC conditions, Zth is equal to Rth.
Equation (4) represents the temperature rise for a particular power dissipation:
ΔTj = |Zth(j - mb)| x Power					[4]A worked example is discussed in the appendix (Section 3.1.2).

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Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

2.6 Electrical characteristics
This section is used to determine whether the MOSFET would be suitable in aparticular application. This section differs from the previous two sections that areused to determine whether the MOSFET would survive within the application. The
examples in this section are taken from the data sheet for the BUK7Y3R5-40H
unless otherwise stated.

2.6.1 Static characteristics
The static characteristics are the first set of parameters listed in this section and an
example is shown in Table 5:

Table 5: Static characteristics
Symbol Parameter

V(BR)DSS

VGS(th)

IDSS

IGSS

RDS(on)RG

38drain-source
breakdown
voltagegate-source
thresholdvoltagedrain leakage
current

gate leakage
current

drain-source
on-state
resistanceConditions

Min

Typ

ID = 250 µA; VGS = 0 V; Tj = 25 °C

4042.7 -

V

ID = 250 µA; VGS = 0 V; Tj = -40 °C-

40.1 -

V

ID = 250 µA; VGS = 0 V; Tj = -55 °C

3639.7 -

V

2.4

3

3.6

V

-

-

4.3

V

1

-

-

V

VDS = 40 V; VGS = 0 V; Tj = 25 °C-

0.03 1µA

VDS = 16 V; VGS = 0 V; Tj = 125 °C

-

1

10

µA

VDS = 40 V; V GS = 0 V; Tj = 175 °C

-

37

500

µA

VGS = 20 V; VDS = 0 V; Tj = 25 °C

-

2

100

nA

VGS = -10 V; VDS = 0 V; Tj = 25 °C

-

2

100

nA

VGS = 10 V; ID = 25 A; Tj = 25 °C

2

2.9

3.5

mΩ

2.7

4.1

5.2

mΩ

2.9

4.5

5.6

mΩ

3.4

5.4

6.7

mΩ

2

Ω

ID = 1 mA; VDS =VGS ; Tj = 25 °C;
Figure 6; Figure
ID = 1 mA; VDS =VGS ; Tj = -55 °C;
Figure 6
ID = 1 mA; VDS =VGS ; Tjj = 175 °C;
Figure6

VGS = 10 V; ID = 25 A; Tj = 105 °C;
Figure9
VGS = 10 V; ID = 25 A; Tj = 125 °C;
Figure9
VGS = 10 V; ID = 25 A; Tj = 175 °C;
Figure9gate resistance f = 1 MHz; Tj = 25 °C

0.32 0.8

Max Unit• VBR(DSS) (drain-source breakdown voltage) - an expansion of the parameter listed
and explained in Section 2.4. This section lists the minimum voltage the device isguaranteed to block between the drain and source terminals in the off-state overthe entire MOSFET temperature range. The temperature range is from -55 °C to+175 °C. The current between the drain and the source terminals of the BUK7Y3R540H when testing V(BR)DSS is 250 µA at the temperatures stated. V(BR)DSS is 40 V orless if the device is cooler than +25 °C and 40 V if the device is between +25 °C and+175 °C.
The effect of temperature on the off-state characteristics is twofold. The leakage
current increases with temperature, turning the device on. Competing against theleakage current increase, the breakdown voltage also increases with temperature.
• VGS(th) (gate-source threshold voltage) is important for determining the on-state
and the off- state of the MOSFET. VGS(th) is defined where VDS = VGS, although it issometimes quoted for a fixed VDS (e.g. 10 V). Note that the definition of thethreshold voltage for a particular current where the gate and drain are shorted
together, can differ from examples in textbooks. The parameter in textbooks
describes a change in the physical state of the MOSFET and is independent of the
MOSFET chip size. The parameter used in the data sheet is for a specified current
and is dependent on the chip size, as the current flow is proportional to the chiparea.
The threshold voltage in the data sheet is defined in a way that is best for routine
measurement, but not how the actual device would typically be used.
Consequently, the graphs provided in Figure 6 and Figure 7.

Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

10-1
ID
(A)

5
VGS(th)
(V)

10-2

4
Max
Typ

3

Min

10-3

Typ

Max

10-4

2
Min

10-5

1

0
-60

-30

0

30

60

ID = 1 mA; VDS = VGS

90

120 150
Tj (°C)180

aaa-018139

10-6

0

1

2

3

4
VGS (V)Tj = 25 °C; VDS = 5 V

5aaa-018138

Figure 6 | Gate-source threshold voltage as a

Figure 7 | Subthreshold drain current as afunction of junction temperature 			function of gate-source voltage

Figure 6 shows the variation in the threshold voltage for the typical and limit
devices over the rated temperature range. All the MOSFETs are guaranteed to havea threshold voltage between the lines.

39Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

Consequently, for the BUK7Y3R5-40H at 65 °C, if VDS and VGS are both less than 2 V,
all devices carry less than 1 mA. Also, all devices carry more than 1 mA if VDS and VGS
are both greater than 4 V. At 175 °C, the lower limit has fallen to 1 V, while theupper limit has fallen to 2.4 V. The lower limit is usually more important as it
determines when the device is guaranteed to be turned off, and the gate driver
performance an application needs.
Figure 7 shows how the device turns on around this threshold voltage. For the
BUK7Y3R5-40H, the current increases 100,000 times for an increase in gate voltageof less than 1 V. An example is given for the situation when the drain-source
voltage is fixed at 5 V.
• IDSS (drain leakage current) guarantees the maximum leakage current that thedevice passes at its maximum rated drain-source voltage(40V in this case) during
the off-state. It is important to note how much higher IDSS is at high temperature,
which is the worst case.
• IGSS (gate leakage current) guarantees the maximum leakage current throughthe gate of the MOSFET. The IGSS is important when calculating how much current isrequired to keep the device turned on. Because it is a leakage current through an
insulator, this current is independent of temperature, unlike IDSS.
• RDS(on) (drain-source on-state resistance) is one of the most importantparameters. The previous parameters guarantee how the device functions when it
is off, how it turns off and what leakage currents could be expected. These factors
are important when battery capacity is an issue in the application.
RDS(on) is a measure of how good a closed-switch the MOSFET is, when turned-on. It
is a key factor in determining the power loss and efficiency of a circuit containing a
MOSFET. The on-resistance RDS(on) x ID2 gives the power dissipated in the MOSFET
when it is turned fully on. Power MOSFETs are capable of carrying tens or
hundreds of amps in the on-state.
Power dissipated in the MOSFET makes the die temperature rise above that of its
mounting base. Also when the MOSFET die temperature increases, its RDS(on)increases proportionally. Maximum recommended junction temperature is 175 °C(for all Nexperia packaged MOSFETs).
Using the BUK7Y3R5-40H data sheet as an example:
Rth(j-mb) temperature rise per Watt between junction (die) and mounting base =1.3 K/W (1.3 °C/W).
Maximum power dissipation for temperature rise of 150 K (Tmb = 25 °C, Tj = 175 °C)= 150 / 1.3 = 115.38 W, note data sheet Ptot is rounded to 115 W.
Maximum RDS(on) at a die temperature (Tj) of 175 °C = 6.7 mΩ.
Therefore, at steady state with Tmb = 25 °C and Tj = 175 °C; P = 115.38 W = ID2 ×40

RDS(on)(175 °C). Therefore:						

[5]

= 131.23 A (rounded down to 120 A in the data sheet).
Note this value appears on the curve for ID vs Tmb in Figure 1 but only 120 A isclaimed in the data sheet. Practically, limitations are placed on MOSFET
performance due to PCB, thermal design and operating temperature, which will all
act together to raise the mounting base temperature.
Figure 8 shows the dependency of RDS(on) on the gate-source voltage for a standard
level MOSFET (e.g. BUK7Y12-55B); the red dashed line shows the curve for a hot
device and is indicative of how the dependency changes at a high temperature.
35
RDSon
(mΩ)
28

21
Hot RDSon

14

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1

MOSFET & GaN FET Application Handbook

7

0

4

8

12

16

VGS (V)20
aaa-002470

This diagram is for illustrative purposes only and not to be taken as an indication of hot RDS(on)performance for any device.
Figure 8 | Drain-source on-state resistance as a function of gate-source voltage at 25 °C and hightemperature

If an application requires good RDS(on) performance for lower gate-source voltages,
then MOSFETs are made with lower threshold voltages, e.g. the BUK9Y12-55B(logic level MOSFET). However, the lower threshold voltage of such a device meansthat it has a lower headroom for its off-state at high temperature. This lower
headroom often means that a device with a higher threshold voltage is needed.

Figure 9 shows how the ratio of RDS(on) to RDS(on) at 25 °C typically varies withjunction temperature for BUK7Y3R5-40H.

41

Understanding power MOSFET data sheet parameters

142

Nexperia | A Power Design Engineer's Guide

2
a
1.6

1.2

0.8

0.4

0
-60

-30

0

30

60

90

RDSon
a−
=
RDSon 25 °C

120 150
Tj (°C)180

aaa-018451

Figure 9 | Normalized drain-source on-state resistance factor as a function of junction temperature

RG (gate resistance) is the internal series resistance presented to the gate pin.
In most applications an external “gate stopper” resistance of a larger value isemployed to mitigate gate oscillation and, dependent on value chosen, decrease
output slew rate.

2.6.2 Dynamic characteristics
The dynamic characteristics determine the switching performance of the device.
Several of these parameters are highly dependent on the measurement conditions.
Consequently, understand the dynamic characteristics before comparing data
sheets from suppliers with different standard conditions. Table 6 is a sample
dynamic characteristics table.

Table 6: Dynamic characteristics

List of constants and limitations relating to the table i.e. voltages, currents and temperatures

Symbol Parameter

QGD

total gatechargegate-source
chargegate-drainchargeCiss

input
capacitance

QG(tot)
QGS

Coss
Crss

outputcapacitancereversetransfer
capacitanceConditions

Min

Typ

Max Unit

ID = 25 A; VDS = 32 V; VGS = 10 V;

-

31

53

nC

-

10

15

nC

-

6

15

nC

-

2294 3441 pF

-

682

954

pF

-

112

247

pF

-

10

-

ns

VDS = 30 V; RL = 1.2 Ω; VGS = 10 V; RG(ext)
=5Ω
-

8

-

ns

VDS = 25 V; VGS = 0 V; f = 1 MHz; Tj= 25 °C;td(on)

turn-on delay
time

tr

rise timetd(off)turn-off delay
time

-

19

-

ns

tf

fall time

-

9

-

ns

-

0.8

1.2

V

-

25

-

ns

-

16

-

nC

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MOSFET & GaN FET Application Handbook

Source-drain diode
VSD
trr
Qr

source-drain
voltage

IS = 25 A; VGS = 0 V; Tj = 25 °C;

reverserecovery time IS = 25 A; dIS/dt = -100 V/μs; VGS= 20 V;
recoveredcharge43Understanding power MOSFET data sheet parameters

1

Nexperia | A Power Design Engineer's Guide

2.6.2.1 Gate charge
QG(tot), QGS, and QGD are all parameters from the same gate charge curve. They
describe how much gate charge the MOSFET requires to switch, for certain
conditions. This is particularly important in high frequency switching applications.
Much of the power loss occurs during switching, when there are significant voltage
and current changes simultaneously between the drain, gate and source. In theblocking state, there are significant voltages but negligible currents. In the full-on
state, there are significant currents and small voltages.
The gate charge parameters are dependent on the threshold voltage and theswitching dynamics as well as the load that is being switched. There is a difference
between a resistive load and an inductive load.
An example of a gate charge curve is shown in Figure 10:
VG
QG(tot)QGS (plateau)

QGD
VGS(th)VD
QGS(th)QGS (plateau)

QG(tot)ID
QGS (plateau)

QGS(th)aaa-002471

Figure 10 | Gate charge curve also showing drain-source currents and voltages

Due to capacitance variation with voltage and current, it is better to look at thegate charge data rather than the capacitance data when determining switching44

performance. This is especially true if the gate-driver circuit for the MOSFET islimited to a particular current, and a rapid switch is required
The gate charge curve describes what happens to a MOSFET which has a drain
supply limited to a particular current and voltage. The operation of the test circuitmeans that during the gate charge curve, the MOSFET is provided with either aconstant voltage or a constant current.
During this time, the drain-source voltage begins to fall because the increased
charge on the MOSFET allows easier conduction. Consequently, although thegate-source voltage is constant, the drain-gate voltage is falling.
Eventually the capacitance stops increasing and any further increases in gatecharge increase the gate-source voltage. This characteristic is sometimes referred
to as the "Miller plateau" as it refers to the time during which the so-called Miller
capacitance increases. The Miller plateau is also known as the gate-drain charge(QGD).
During this period, there are significant currents and voltages between the drain
and source, so QGD is important when determining switching losses.
Once the end of the Miller plateau is reached, the gate-source voltage increases
again, but with a larger capacitance than before QGS had been reached. The
gradient of the gate charge curve is less above the Miller plateau.

1
Understanding power MOSFET data sheet parameters

MOSFET & GaN FET Application Handbook

The gate-charge parameters are highly dependent on the measurement conditions.
Different suppliers often quote their gate-charge parameters for different
conditions, demanding care when comparing gate charge parameters fromdifferent sources.
Higher currents lead to higher values of gate-source charge because the plateau
voltage is also higher. Higher drain-source voltages, lead to higher values ofgate-drain charge and total gate charge, as the plateau increases.
The drain-source currents and voltages during the gate charge switching period areshown in Figure 10.
If the MOSFET starts in the off state (VGS = 0 V), an increase in charge on the gate
initially leads to an increase in the gate-source voltage. In this mode, a constantvoltage (VDS) is supplied between the source and drain.
When the gate-source voltage reaches the threshold voltage for the limiting
current at that drain- source voltage, the capacitance of the MOSFET increases andthe gate-voltage stays constant. This is known as the plateau voltage and the onset
charge is referred to as QGS. The higher the current is, the higher the plateau

45Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

voltage (see Figure 11). This relates to the transfer characteristic, see Figure 15.
The transfer curve shows the dependency of drain current on gate voltage. The
higher the voltage on the gate the higher the charge applied, (Q = C × V), hence
making it the easier for the MOSFET to conduct.

VGS
(V)

reduced VDS
QGS

Qtot

QGD

reduced IDS

Vplateau

QG (nC)

aaa-002472

Figure 11 | Features of gate charge curve

2.6.2.2 Capacitances
Capacitance characteristics are generally less useful than the gate chargeparameters, for the reasons already discussed. However, they are still listed on data
sheets. The three capacitances that are normally listed are as follows:• Ciss (input capacitance) is the capacitance between the gate and the other two
terminals (source and drain).
• Coss (output capacitance) is the capacitance between the drain and the other two
terminals (gate and source).
• Crss (reverse transfer capacitance) is the capacitance between the drain and thegate. See Figure 13.
Semiconductor capacitances generally depend on both voltage and the frequencyof the capacitance measurement. Although it is difficult to compare capacitances
measured under different conditions, many suppliers specify a measurement
frequency of 1 MHz. Consequently, the capacitances vary with drain-source voltage(see Figure 12). However, the capacitances also vary with gate-source voltage,
which is why the gradients in the gate-charge curve vary for different voltages (seeFigure 10).
The relationship between charge, voltage and capacitance in the gate charge curve46is: ΔQ = ΔC x ΔV. For different gradients at different gate voltages, the capacitancechanges significantly with gate-source voltage.
104
C
(pF)

CGD
NMOS

Ciss

CDS

103

CGS

Coss

10
10-1

Ciss = CGS + CGD
Coss = CDS + CGD
Crss = CGD

Crss
rss

102

1

10

VGS = 0 V; f = 1 MHz

VDS (V)102

aaa-026238

aaa-02233

Figure 12 | Capacitances as a function of drain source

Figure 13 | MOSFET capacitancevoltage					parameters

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MOSFET & GaN FET Application Handbook

2.6.2.3 Switching times
Most manufacturers quote resistive load switching times. However, extreme care isneeded when comparing data from different manufacturers, as they are highly
dependent on the resistance of the gate drive circuit used for the test and in thecase of logic level devices, the gate voltage applied.
In devices for fast switching applications, the gate resistance of the MOSFET isoften quoted as capacitive time constants which are equally dependent on
resistance and capacitance.

2.6.2.4 Diode characteristics
The diode characteristics are important if the MOSFET is being used in the so-called"third quadrant". The third quadrant is a typical arrangement where the MOSFET
replaces a diode to reduce the voltage drop from the inherent diode forward
voltage drop. In such a situation, there is always a small time period when the
MOSFET parasitic diode is conducting before the MOSFET turns on. For suchapplications, the diode switching parameters are important. In addition, diodereverse recovery (Figure 14) contributes to the power losses as well as oscillationand voltage spikes, which can cause EMC concerns.

47

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

ID
(A)

trr
dI/dt

ta

tb

0

0.25 IRM
dIR/dtIRM

trr = ta + tbt (s)Figure 14 | Reverse recovery waveform definitions

2.6.2.5 Transfer characteristics
Figure 15 shows the transfer characteristics of the MOSFET which indicate thetheoretical drain current that can be handled by the device as a result of applied
gate to source voltage. The graph shows the temperature dependency of thecharacteristic. The convergent point indicates where the current is dependent only
on gate voltage and not temperature hence the MOSFET’s Zero Temperature
Coefficient point I_ZTC. Gate charge test current and plateau voltage (refer toFigure 10) can be directly read from the transfer curve.
200
ID
(A)
160

120

80
175ºC
40
25°C
0

Tj = -55°C
01

VDS = 12 V

2

3

4

5
VGS (V)6
aaa-026234

Figure 15 | Transfer characteristics; drain current as a function of gate-source voltage; typical values48

3 Appendices3.1 Safe Operating Area (SOA) curves
To highlight the key features, Figure 16 provides an idealized SOA curve for ahypothetical MOSFET. Data for a hypothetical MOSFET for a single pulse length, isshown to highlight the region where it deviates from the ideal curve.
102

(2)

IDS
(A)

(3)
(1)

10

(5)

(4)

1

10-1

1.
2.
3.
4.
5.

1

10

102

VDS (V)

103

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MOSFET & GaN FET Application Handbook

RDS(on) limit (VDS/IDS is constant)
Maximum pulsed drain current (IDS is constant)
Maximum pulsed power dissipation (VDS x IDS is constant)
Maximum allowed voltage (VDS is constant)Linear mode derating - a departure from the ideal behavior shown in (3) due tooperation within a regime of positive feedback, and potential thermal runaway
aaa-002473

Figure 16 | Idealized SOA curve at a single time pulse for hypothetical MOSFETThe dashed line (5) is to emphasize where the curve deviates from the ideal. In
reality, there is a single curve with a change of gradient where the linear mode
derating becomes important.

49

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

RDS(on) limit
RDS(on) is region (1) of the graph and Equation [6] represents the limiting line:
VDS_____ ≤ R

DS(on)(max) (175ºC)

				

[6]

IDS
The limit is when the MOSFET is fully on and acting as a closed switch with aresistance that is no greater than the hot RDS(on).
Constant current region
The constant current region is region (2) of the graph. It is the maximum pulsed
drain current, which is limited by the device manufacturer (for example, thewire-bonds within the package).
Maximum power dissipation (linear mode) limit
In this region, the MOSFET is acting as a (gate) voltage-controlled current source.
This means that there are significant voltages and currents applied simultaneously,
leading to significant powerdissipation. Line (3) shows the idealized curve, whereas
the dotted line (5) shows where it deviates from the ideal.
The limiting factor for the SOA curve in region (5), is the heating applied during a
rectangular current and voltage pulse. Even in the ideal situation, this curve
depends on the transient thermal impedance of the MOSFET, which is covered in
Section 2.5.
The transient thermal impedance varies with the pulse length. This is due to thedifferent materials in the MOSFET having different thermal resistances andcapacities. The differences create a thermal equivalent to an RC network from thejunction (where the heat is generated) to the mounting base. Equation (7) is thecalculation used to determine the ideal curve in this region.

P = ID x VDS

Tj(max) _ Tmb_________== Constant

			

[7]

Zth(j-mb)

The ideal situation accurately describes the situation for sufficiently high current
densities. However, it is overly optimistic for low current-densities, i.e. towards thebottom right of region (3). Low current densities and high voltages can lead tothermal runaway in the linear mode operation. Thermal runaway is discussed in thefollowing section.

50

Thermal runaway in linear mode
Power MOSFETs are often considered to be immune to thermal runaway due to thetemperature coefficient of resistance, which means that as temperature rises,
current falls.
This is only true for MOSFETs that are fully on (i.e. in region 1), but it is not thewhole story.
When a MOSFET is turned on, there are two competing effects that determine how
its current behaves with increasing temperature. As the temperature rises, thethreshold voltage falls. The MOSFET is effectively turned on more strongly, thereby
increasing the current. In opposition, the resistance of the silicon increases withincreasing temperature, thereby reducing the current. The resultant effect for aconstant drain-source voltage, is shown in Figure 17. This situation occurs when thegate-source voltage of a MOSFET is being used to control the current, or when the
MOSFET is switched sufficiently slowly.

ID
(A)

25 °C
negative temp.
coefficient

175 °C

Understanding power MOSFET data sheet parameters

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MOSFET & GaN FET Application Handbook

ZTC point

positive temp.
coefficient

VGS (V)
aaa-02474

Figure 17 | Transfer characteristics for a hypothetical MOSFET, showing regions of positive andnegative temperature coefficient

The resistance increase dominates at high currents, meaning that localized heating
leads to lower currents. The threshold-voltage drop dominates at low currents,
meaning that localized heating lowers the threshold voltage. This condition
effectively turns on the device more, leading to higher currents and a risk ofthermal runaway.
Consequently, for a given VDS, there is a critical current below which there is apositive-feedback regime and a subsequent risk of thermal runaway. Above thiscritical current, there is negative feedback and thermal stability. This critical current

51

Understanding power MOSFET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

is known as the Zero Temperature Coefficient (ZTC) point.
This effect reduces the SOA performance for low currents and high drain-source
voltages. The constant power line must be reduced as shown in region (5). For
short switching events, this effect is insignificant. However, as the duration of theswitching event becomes longer, for example to reduce electromagnetic
interference, the effect becomes more important and potentially hazardous.
Voltage-limited region
The device is limited by its breakdown voltage VDS which is shown in region (4). The
quick reference data provides values for VDS at temperatures of 25 °C and above. In
the hypothetical MOSFET shown in Figure 16, the rating is 100 V. For the
BUK7Y3R5-40H, the voltage is 40 V.

3.1.1 Safe operating area for temperatures above 25 °C
When reading from the SOA curves there are two main considerations:
1. The mounting base temperature is at 25 °C
2. The MOSFET is exposed to a rectangular power pulse
The SOA curves indicate the pulse power required to raise the MOSFET junction
temperature, Tj, from 25 °C to its maximum rating of 175 °C, Tj(max).The assumption
being Tj is at the mounting base temperature Tmb = 25 °C when power dissipation
begins.
Under DC conditions, where a product has been operating for some time (thermallysoaked), Tj will be close to the mounting base temperature, Tmb.
Under high power short duration pulse conditions Tj, can be much greater than Tmb.
A reduction in MOSFET power handling capability results if the initial Tmb > 25 °C, so
it is important to establish an accurate temperature for Tmb to determine Tj which is
non-trivial activity.
The majority of automotive power applications thermally link the MOSFET
mounting base via the PCB footprint and an electrically isolating barrier to ametallic mass (e.g. product housing) to act as a heatsink. Heatsink temperature Ths
is in turn thermally linked to the product’s ambient temperature, Tamb, which may
be 85 °C for in-cabin (inside the driver compartment) or 105 °C for under the hood(near and around the engine). In some automotive applications where, for example,
the housing is mounted directly to the power train the customer may specify atemperature directly on the heatsink where Ths ≥ 105 °C.

52

Unless the offset relationship Tmb > Tamb is well defined by thermal flow modelling
and test verification, the designer can only estimate Tmb.
It is important to remember that in a product which has been operating for sometime (thermally soaked) MOSFETs which have similar thermal linkages to the sameheatsink will have a similar Tmb, whether the MOSFET is switched on or switchedoff.
Calculation to determine the rise in die temperature above mounting basetemperature is given below, see Equation (8).
Tj(rise) = Tj(max) − Tj(mb) = Ppeak . Zth(t)

				

[8]

Where:
Ppeak is the peak power, as this is a rectangular pulsed waveform.
Zth(t) is the thermal impedance value between junction and mounting base for apulse of t seconds duration.
Note: In DC applications, (where the MOSFET is not pulsed), use Rth(j-mb) instead ofZth(j-mb). Generally, for pulses above 100 ms, Zth is indistinguishable from Rth.
Equation 8 assumes heating is uniformly spread across the whole die. A
combination of higher VDS with lower IDS and pulse durations ≥ 100 μs can cause
hot spotting (uneven heating) to occur known as the Spirito effect. Refer toTN00008 Section 5 (chapter 13 in this book) - Power and small-signal MOSFET
frequently asked questions and answers.

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MOSFET & GaN FET Application Handbook

3.1.1.1 Example calculations
Calculate the max DC IDS for BUK7Y3R5-40H, with VDS = 10 V at 25 °C
Rewrite Equation (10) to bring out IDS as the main subject (Power is substituted by
IDS × VDS). Since Equation (10) is being used for a DC situation, the Zth(t) parameterused in Equation (8) and Equation (9) is replaced by the Rth steady state condition.
Tj(rise) = IDS × VDS × Zth(t) Tj(rise)
Tj(rise)________________

= IDS

				

[9]

					 [10]VDS x Rth

175 ºC - 25 ºC________________= 11.3 A

			

[11]

10 V x 1.3 K/W

53

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Nexperia | A Power Design Engineer's Guide

When making these calculations always refer to the Safe Operating Area Curve forthe device. The SOA curve for the BUK7Y3R5-40H is Figure 4 which can be found in
Section 2.4.4. The intersection of ID = 11.3 A and VDS = 10 V occurs above the DC
curve and therefore this condition exceeds device capability. The reason being heat
distribution is non-uniform within the die. In other words hot spotting is present,
otherwise known as the Spirito effect.
The BUK7Y3R5-40H SOA curve shows for a DC condition the Spirito effect for thisdevice (the point at which a gradient change occurs) is evident at VDS = 0.94 V for
ID= 120 A. As ID reduces, the allowable VDS increases and by ID = 11.3 A the
maximum permissible VDS = 4.4 V. Thermal consideration are discussed further in
Section 3.1.2.

3.1.2 Example using the SOA curve and thermal characteristics
Consider the following application during linear mode operation:• Ipulse = 20 A
• Vpulse = 30 V
• f = 2 kHz• tpulse = 100 μs• Tamb = 25 °C

3.1.2.1 Calculation steps
The SOA curve (Figure 4) is initially checked to see whether any single pulse would
cause a problem. Observing the SOA curve, it can be seen that the 20 A, 30 V pulse
lies between the 100 μs and 1 ms lines. This indicates that the pulse lies within
acceptable limits.
The duty cycle for the pulses is now calculated using a frequency of 2 kHz for
100 μs pulses. These values give a duty cycle of 0.2. The Zth curve (Figure 5)
indicates that for 100 μs, the line with the duty cycle (δ) has a transient thermal
impedance of 0.35 K/W.
The power dissipation for the square pulse is 20 A × 30 V, which equals 600 W.
Using Equation (8), the temperature rise for the 100 μs pulse is calculated as being
600 W × 0.35 K/W, which equals 210 K. With a starting temperature of 25 °C, thetemperature rise results in a finishing temperature of 235 °C. As the MOSFET
junction temperature must not exceed 175 °C, the MOSFET is not suitable for thisapplication.
If the application requires a single pulse, then the curve shows that the transient
thermal impedance for a 100 μs pulse is 0.12 K/W. As a result, the temperature riseis 600 W × 0.12 K/W which equals 72 K. The finishing temperature is then 97 °C for54

a starting temperature of 25 °C. The device is able to withstand this, thereby
confirming what the SOA curve already indicated.

3.1.2.2 Derating for higher starting temperatures
The example Safe Operating Area calculations were performed for a mounting
base temperature of 25 °C. At higher mounting base temperatures, the SOA curves
must be de-rated, as the allowed temperature rise in the junction is reduced. The
allowed power of the pulse is reduced proportionally to the reduced temperaturerise. For example, with a mounting base temperature of 25 °C, the allowed
temperature rise is 150 °C. At 100 °C, the allowed temperature rise is half of that
(75 °C). The allowed power is half of that allowed at 25 °C.
The allowed drain current is de-rated while the allowed drain-source voltage ismaintained.
The ID derating of the SOA curve at 100 °C is shown in Figure 18.
103
ID
(A)
30 A at 100 °C

8 V, 30 A at Tmb = 100 °C
is above the 1 ms curve - outside SOA

Limit RDS(on) = VDS / ID
102

tp = 10 us

10Understanding power MOSFET data sheet parameters

1

MOSFET & GaN FET Application Handbook

DC
0.5 A at 100 °C

8 V, 30 A at Tmb = 25 °C
is below the 1 ms curve - inside the SOA

1

1 ms
10 ms
100 ms0.05 A at 100 °C

10-1
10-1

1

10

VDS (V)102

Tmb = 25 °C; IDM is a single pulse
aaa-02475

Figure 18 | SOA curve showing derating from Tmb = 25 °C to Tmb = 100 °C

Example:Is a 1 ms pulse of 30 A and 8 V allowed at 100 °C?
See Figure 18, the intersection of the blue lines of the rectangular power pulse ID =30 A * VDS = 8 V below the 1 ms curve indicates that the device can safely handle
this pulse with Tmb = 25 °C .
Power handling capability halves with Tmb = 100 °C. In Figure 18 the red text shows
ID at 100 °C is half the value at 25 °C.
The intersection of the red lines of the rectangular power pulse ID = 30 A * VDS = 8 V
above the 1 ms curve indicates the device cannot handle this pulse at Tmb = 100 °C.

55

Understanding power MOSFET data sheet parameters

1

56

Nexperia | A Power Design Engineer's Guide

4 References
1. The Impact of Trench Depth on the Reliability of Repetitively Avalanched
Low-Voltage Discrete Power Trench nMOSFETs - Alatise et al, IEE Electron Device
Letters, Volume 31, No7, July 2010, pages 713-715.
2. Semiconductor Devices - Physics and Technology S.M.Sze, 1985, John Wiley andSons.
3. Nexperia application note AN10273 - Power MOSFET single-shot and repetitive
avalanche ruggedness rating.
4. Nexperia application note AN10874 - LFPAK MOSFET thermal design guide.
5. Nexperia application note AN90001 - Designing in MOSFETs for safe and reliablegate-drive operation.
6. Nexperia technical note TN00008 - Power and small-signal MOSFET frequently
asked questions and answerss

5 Tables
Table 1. Quick reference data............................................................................................ 27
Table 2. Pinning information.............................................................................................. 29
Table 3. Limiting values....................................................................................................... 30
Table 4. Thermal characteristics........................................................................................ 37
Table 5. Static characteristics............................................................................................. 38
Table 6. Dynamic characteristics ...................................................................................... 43

1
Understanding power MOSFET data sheet parameters

MOSFET & GaN FET Application Handbook57

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158

Nexperia | A Power Design Engineer's Guide

6 Figures
Figure 1. Continuous drain current as a function of mounting base		
temperature.......................................................................................................33
Figure 2. Normalized total power dissipation as a function of mounting
		base temperature..............................................................................................34
Figure 3. Avalanche current as a function of avalanche period.................................35
Figure 4. Safe operating area; continuous and peak drain currents as a		function of drain-source voltage....................................................................36
Figure 5. Transient thermal impedance from junction to mounting base		as a function of pulse duration........................................................................37
Figure 6. Gate-source threshold voltage as a function of junction		
temperature.......................................................................................................39
Figure 7. Subthreshold drain current as a function of gate-source voltage............39
Figure 8. Drain-source on-state resistance as a function of gate-source		voltage at 25 °C and high temperature.........................................................41
Figure 9. Normalized drain-source on-state resistance factor as a function
		of junction temperature...................................................................................42
Figure 10. Gate charge curve also showing drain-source currents and voltages......44
Figure 11. Features of gate charge curve.........................................................................46
Figure 12. Capacitances as a function of drain source voltage..................................47
Figure 13. MOSFET capacitance parameters...................................................................47
Figure 14. Reverse recovery waveform definitions........................................................48
Figure 15. Transfer characteristics; drain current as a function of gate-source ...........
voltage; typical values.......................................................................................48
		Figure 16. Idealized SOA curve at a single time pulse for hypothetical MOSFET......49
Figure 17. Transfer characteristics for a hypothetical MOSFET, showing		regions of positive and negative temperature coefficient........................51
Figure 18. SOA curve showing derating from Tmb = 25 °C to Tmb = 100 °C................55

Chapter 2
Designing in MOSFETs
for safe and reliablegate-drive operation
Application Note: AN90001

Designing in MOSFETs for safe and reliable gate-drive operation

2

MOSFET & GaN FET Application Handbook59

Designing in MOSFETs for safe and reliable gate-drive operation

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Nexperia | A Power Design Engineer's Guide

The MOSFET gate-source threshold voltage VGS(th) and maximum gate-source
voltage (VGS-max) are key parameters that are critical to the reliable operation of
MOSFETs. The threshold voltage represents the voltage at which the MOSFET
starts to turn on, whilst the maximum gate-source voltage is the maximum
gate-source voltage that the MOSFET can withstand safely. VGS-max ratings vary
between suppliers and between MOSFETs, which can make it difficult to choose
appropriate MOSFETs for the application. This chapter aims to provide the designerwith enough knowledge to appreciate these differences and to select anappropriate MOSFET. We also present a new methodology demonstrating where
higher VGS-max ratings voltages may be applied for Nexperia’s automotive grade
MOSFETs.

1 Introduction
The MOSFET can be considered as a voltage-controlled switch. Applying a voltage(VGS) across the gate and source terminals enhances the MOSFET allowing current
to flow through the MOSFET-channel between the drain and source terminals.
VGS(th) is defined as the VGS for a pre-defined drain current, commonly 1 mA.
Increasing the VGS further causes the MOSFET to become fully enhanced. This
allows more current to flow for a given drain-source voltage. A fully enhanced
MOSFET will also have achieved close to its rated on state resistance RDS(on).
Increasing the voltage beyond the full enhancement level will only result in a small
reduction in RDS(on). Increasing the VGS further to a level beyond the VGS-max will havedirect implications for the MOSFET’s reliability. This could lead to destruction of the
MOSFET. The design engineer can review the MOSFET data sheet to help ensurethat the MOSFET’s reliability is not impaired and that it operates
as expected. The data sheet gives limits and guidelines, but often the design
engineer needs more detail to select a MOSFET.
This chapter aims to bridge this gap. It will provide an overview of what is currently
included in the data sheet concerning VGS ratings and it aims to answers questions
that may arise when designing in power MOSFETs.
In addition a brief introduction to the physics behind the VGS ratings and the role of
the gate-oxide is included. This will provide an appreciation of the reasoning behind
the ratings and give confidence in the methodology employed by Nexperia to test,
to qualify and to rate the MOSFET’s gate-oxide capability.
An analysis of common Power MOSFET applications is also presented. This will
highlight the demands imposed on the MOSFET gate-oxide by the application.
Different applications have been analysed to develop a mission profile for the VGS.
This states what levels of VGS the MOSFET will be exposed to, for how long and at
what temperature. This information can then be used to select the correct MOSFET
for the application, thus ensuring optimum and reliable operation.

60

2 Data sheet VGS ratings
In Nexperia data sheets there are typically two main parameters concerning thegate- source voltages. The first being the VGS threshold value or VGS(th). The second
is the VGS rating, which in this application note will be called VGS-max.
Some MOSFET data sheets include a value for the plateau (Miller) voltage. This is
the VGS voltage during switching events. Its level is poorly defined as it depends on
several interdependent electrical and environmental factors. It is not important for
MOSFET selection or circuit design and will not be further discussed in this chapter.

2.1 VGS threshold voltage - VGS(th)Figure 1 shows an example of data sheet VGS(th) values; it highlights the voltage
required across the gate and source terminals to start to turn the MOSFET on. The
conditions for which the VGS(th) is defined are specified here as the VGS necessary for
1 mA of current to flow through the drain terminal. The Nexperia data sheet gives
the VGS(th) for a range of temperatures. It also provides minimum and maximum
values to account for process variation.
The VGS(th) is the start of MOSFET enhancement, an increase in VGS is required toenhance the MOSFET further. Depending on whether a device is logic-level orstandard- level, a MOSFET can be considered fully-enhanced (or fully on) when the
VGS is 5 V or 10 V respectively. By this point the MOSFET has achieved its rated
RDS(on).

Symbol Parameter

VGS(th)gate-sourced
thresholdvoltage

Conditions

Min

Typ

Max

Unit

ID = 1 mA; VDS = VGS; Tj = 25 ºC;
Figure 9; Figure 10

2.4

3

4

V

ID = 1 m; VDS = VGS; Tj = -55 ºC;
Figure 9

-

-

4.5

V

ID = 1 mA; VDS = VGS; Tj = 175 ºC;
1Figure 9--

V

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

Figure 1 | VGS threshold values for a Nexperia automotive grade standard level device

61Designing in MOSFETs for safe and reliable gate-drive operation

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Nexperia | A Power Design Engineer's Guide

Symbol Parameter

VGS(th)gate-sourced
thresholdvoltage

Conditions

Min

Typ

Max

ID = 1 mA; VDS = VGS; Tj = 25 ºC;
Figure 9; Figure 10

1.4

1.7

2.1

V

ID = 1 m; VDS = VGS; Tj = -55 ºC;
Figure 9

-

-

2.45

V

-

-

V

ID = 1 mA; VDS = VGS; Tj = 175 ºC;
0.5
Figure 9

Figure 2 | VGS threshold values for a Nexperia automotive grade logic level device

Other differences between standard-level and logic-level devices can be observedin their threshold-voltage values (Figures 1 and 2). It is important to consider thesevalues when selecting a MOSFET for an application. Two questions need to beanswered to choose an appropriate MOSFET.
1. Can the gate-driver turn the MOSFET (fully) on?
2. Can the gate-driver turn the MOSFET (fully) off?
To answer these questions, it is necessary to take the worst case conditions in eachcircumstance. In order to assess question 1 we need to know what is the highest
value of VGS(th) that the device may have. From Figures 1 and 2, we can see thisoccurs at the lower temperature limits. For a standard level device this can be as
high as 4.5 V. If the gate-driver output is only 5 V, then a standard level device may
not be suitable. This is because the MOSFET is only beginning to turn on andtherefore may not operate correctly in the application. A logic level MOSFET with amaximum VGS(th) of ≤ 2.45 V would be more suitable.
It is important to note that VGS(th) has a Negative Temperature Coefficient (NTC): as
the junction temperature increases, the VGS for a given ID decreases. If channel
current flows because the VGS is not low enough to drive the MOSFET fully OFF, ID
will increase as die temperature rises. This thermal runaway effect can ultimately
cause MOSFET destruction.
Conversely, the designer must consider the upper temperature limit for turning the
MOSFET off. At 175 °C a logic level device may conduct channel current at a
VGS ≥ 0.5 V. The gate-driver may have a minimum output voltage of 0.6 V. This is not
low enough to ensure that a logic level MOSFET is turned fully off. At VGS = 0.6 V
channel current could still flow. In this situation it is better to select a standard level
device.

622.2 The maximum VGS ratings - VGS-max
VGS-max is the absolute maximum gate-source voltage that can be applied to thedevice. These values illustrated in Figures 3 and 4 conform to the International
Electrotechnical Commission’s IEC60134 document -“Rating systems for electronic
tubes and valves and analogous semiconductor devices”. Referencing paragraph 4of IEC60134 titled “Absolute maximum rating system” it states [1]:"Absolute maximum ratings are limiting values of operating and environmental
conditions applicable to any electronic device of a specified type as defined by its
published data, which should not be exceeded under the worst probable conditions.
These values are chosen by the device manufacturer to provide acceptable serviceabilityof the device, taking no responsibility for equipment variations, environmental
variations, and the effects of changes in operating conditions due to variations in thecharacteristics of the device under consideration and of all other electronic devices inthe equipment."
This defines the worst case conditions and highlights the responsibility of thedesigner to ensure that the absolute maximum value is never exceeded. It does not
explicitly place any obligations on the manufacturer in terms of long term
reliability.

Symbol Parameter

Conditions

Min

Max

Unit

VGS

Tj ≤ 175 ºC; DC

-20

20

V

gate-sourced voltage

Figure 3 | Maximum VGS rating for a Nexperia automotive grade standard level device

Symbol Parameter
VGS

gate-sourced voltage

Conditions

Min

Max

Unit

Tj ≤ 175 ºC; DC

-10

10

V

-15

15

V

Tj ≤ 175 ºC; Pulsed[1] [2]

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

Notice there is an additional pulsed value limited to an accumulated duration of up
to 50 hours.
Figure 4 | Maximum VGS rating for a Nexperia automotive grade logic level device

63Designing in MOSFETs for safe and reliable gate-drive operation

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3 Life Testing
For automotive MOSFETs, AEC-Q101 qualification is required. AEC-Q101 is a list ofendurance tests the MOSFET must pass. Each test takes a sample of MOSFETs andplaces them under various electrical stresses and temperatures. The sample size is aminimum of 77 MOSFETs; each endurance test is repeated 3 times on a different
sample of 77 MOSFETs to represent a different process batch. Surviving theendurance tests grants the tested product type Q101 accreditation, allowing the
MOSFET to be used in automotive applications [2]. This can be found on all
Nexperia automotive qualifed MOSFET data sheets (Figure 5).

Figure 5 | Front page of a typical Nexperia automotive data sheet, showing the MOSFET isautomotive by highlighting that it is Q101 compliant

To qualify the VGS-max ratings for Q101 the devices undergo a specified life-testprocedure. The devices are placed in a chamber at the maximum rated temperature(Tjmax) with gate biased at 100% of the data sheet maximum voltage rating for1000 hours. These tests are performed by Nexperia to ensure that the data sheet
ratings of VGS-max are within specification.
However, life-testing alone is inherently incapable of determining the actual
capability of a MOSFET in terms of expected failed parts per million (ppm). In orderto satisfy AEC-Q101 criteria no sample devices during the Q101 bias tests should
fail. It is possible to extrapolate a failure rate from such data, but the maximum
confidence it can provide is in the order of 11000 ppm (based on zero fails in a
population of 240 devices under test and using a confidence interval of 95%). With
this level of confidence it is not trivial to determine what to specify as the
maximum voltage rating. Especially when in addition, Nexeria’s own requirements
target a sub 1 ppm defectivity level.

64

To achieve sub-ppm levels Nexperia has developed its own additional testing
methodology. Using this, Nexperia can confidently rate the devices to satisfy the
AEC-Q101 criteria, while also achieving sub 1 ppm defectivity rates (3). In order tounderstand this methodology an explanation of the role of a MOSFET’s gate-oxide
is required and is detailed below.

4 The Gate-oxide
The methodology to determine and qualify the maximum VGS rating of a device isbased upon characterisation of the gate-oxide. This is the layer of oxide that
insulates the gate polysilicon from the source of the MOSFET and is critical to its
operation. Figure 6 shows the location of the gate-oxide in the MOSFET.
Trench Channel

Source Metal

Gate Oxide

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

Gate Polysilicon

Drain

aaa-026548

Figure 6 | Cross-section of a power MOSFET at approximately 25,000 magnification showing where
the MOSFET channel forms and the location of the gate-oxide. The gate-oxide is present both sidesof the trench and a channel forms next to both oxides. Current flows vertically from Drain to Source
through the MOSFET channel.

65Designing in MOSFETs for safe and reliable gate-drive operation

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When designing power MOSFETs, there are two conflicting constraints with respect
to the gate-oxide thickness. Achieving a low RDS(on) at a low applied level of VGS
requires a thin gate-oxide, conversely withstanding high maximum VGS requires a
thick gate-oxide. In most cases, both requirements can be met and gate bias life
testing can be performed at 100 % of the absolute maximum rating. However,there are some instances where this is not possible. Logic level devices aredesigned to provide a guaranteed RDS(on) at VGS = 5 V. This implies a need for agate-oxide that is thinner than for a standard level device. Consequently a logiclevel gate-oxide is not capable of achieving the same maximum VGS-max rating as astandard-level gate-oxide. For Nexperia’s automotive MOSFETs, this is typically 10 V
logic-level and 20 V for standard-level devices at the maximum rated temperature,
typically 175 °C.
However, caution should be exercised when comparing other manufacturers’ data
sheets. The VGS-max rating is not always given at the maximum rated temperature;
instead it is often given for 25 °C. The rating given for a temperature of 25 °C may
be higher than one given for 175 °C. This will make that MOSFET appear morecapable on first review. Because MOSFET gate-oxide reliability decreases withincreasing temperature, Nexperia’s data sheets show the capability of the MOSFET
at maximum rated junction temperature.
Exceeding the VGS-max rating for the gate-oxide does not mean the device will
immediately fail in the application. Applying a VGS to a MOSFET will develop an
electric field across the internal gate-oxide (EOX). The oxide electric field strength isa function of oxide thickness (TOX) and VGS.
EOX = VGS / TOX
If the oxide electric field strength reaches a sufficiently high level, the gate-oxide
insulation will break down and will be destroyed. However, for electric fieldstrengths below this critical value the gate-oxide may degrade over time andultimately fail. The rate at which this occurs depends on both the operatingtemperature and the strength of the electric field within the gate-oxide. Expectedoxide life-time decreases with increases in either temperature or oxide electric-field
stress.
Gate-oxides need to be rated to survive for operating periods of not less than 10
years. Testing for 10 years to assess the reliability of the gate-oxide is not practical.
Nexperia performs life tests under accelerated conditions to achieve quicker test
results. For assessing the gate-oxide, Nexperia places a group of MOSFETs withknown gate-oxide thickness under high temperatures and high oxide field strength.
The MOSFETs will therefore degrade much quicker than can be expected in theapplication. The MOSFETs are monitored until the point of failure, which isobserved as a low impedance measurement between the gate and source
terminals. This is termed Time Dependant Dielectric Breakdown (TDDB).

66

Nexperia uses the test data from TDDB to correlate to other operatingtemperatures and oxide field-strengths to determine the capability of the oxide in
actual application operating conditions. This is applicable to MOSFETs withdifferent gate-oxide thicknesses because the testing characterises oxide
field-strength, not VGS.
A gate-oxide production process aims to create a gate-oxide of consistent thickness
throughout manufacture. However, in any real fabrication process variation will
occur. This means the VGS rating needs to be applicable to the MOSFET with thethinnest gate- oxide that leaves the factory. Production screens are in place that
set a lower limit, which corresponds to the MOSFET with the thinnest gate-oxide.
Using this value of gate-oxide thickness and the characterisation data from theaccelerated TDDB tests, this MOSFET can now have its oxide rated for VGS-max andtemperature for a given expected operating life.
For automotive MOSFET the operating requirement is outlined within AEC-Q101;
1000 hours at VGS-max and maximum temperature. Taking this profile into account,and knowing the thinnest oxide that can be supplied, Nexperia can confidently
place VGS-max ratings in the data sheet.
Again, for further detail on the mechanics of gate-oxide wear out and Nexperia’s
methodology towards rating its gate-oxides please refer to [3].

5 Alternative operating profiles
The mission or operating-profile of a MOSFET in an actual application is rarely at
the data sheet maximum temperature and at the maximum VGS for 1000 hours as
dictated by AEC-Q101. This combination is formulated as a compromise to allow
realistic life-testing times that would protect automotive applications. The ultimate
rating of a MOSFET VGS- max that meets AEC-Q101 may not meet the application
requirements. For example the application may require 12 V VGS but this exceeds
the MOSFET of choice’s 10 V rating, at first pass this MOSFET cannot be used in theapplication. However, if the operating profile of the application is outlined withgreater detail it may be that the 12 V rating is required for temperatures less than
175 °C and for a cumulative lifetime duration of less than 1000 hours. Here theapplication criteria can be compared with the true capability of the oxide andtherefore be confidently selected for the application.

2
Designing in MOSFETs for safe and reliable gate-drive operation

MOSFET & GaN FET Application Handbook

This chapter assesses several common MOSFET topologies in order to provide anunderstanding of VGS behaviour. From this it is possible to generate an estimate forwhat VGS will be applied to the MOSFET and for how long; a VGS profile. Using the
VGS profile along with operating-voltage profiles and temperature profiles a fullmission profile can be created. An example is shown in Table 1. The mission profile
can be used to assess the MOSFET’s suitability for the application against the TDDB

67

Designing in MOSFETs for safe and reliable gate-drive operation

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Nexperia | A Power Design Engineer's Guidedata.
The first set of analyses presented considers steady state stress on the gate-oxide
due to a constant gate-source bias from either an external gate-driver or as a resultof the steady state voltages that appear on the gate and source terminals of a
MOSFET. Further analysis attempts to estimate the contribution to gate stress fromtransient gate- source voltages. This could result from capacitive coupling throughthe Miller capacitance or conduction through the MOSFET body diode. Both requiredetailed analysis of the application to verify compatibility with the MOSFET
features.

Table 1. Example MOSFET mission profile, showing the time spent in hours
at a particular VGS and MOSFET junction temperature
Time (hours)
VGS =Tj = -40 ºC

Tj = 25 ºC

Tj = 80 ºC

Tj = 105 ºC

Tj = 125 ºC

-5 V

1

5

10

1

0

2V

10

50

100

10

2

5V

1000

2500

5000

50

10

10 V

100

200

300

100

5

20 V

2

10

5

2

1

5.1 Application assessments
This section assesses several common topologies featuring MOSFETs and focuseson the VGS behaviour. This information will help to produce the mission profile asshown in Table 1.

5.1.1 Single MOSFETs
Starting with the three application topologies shown in Figure 7. Figure 7a and 7b
can be considered similar in behaviour as they are both driven gate to source.
However, Figure 7c shows the MOSFET being driven gate to ground. It is possible todrive the MOSFET in this manner by ensuring the gate voltage is higher thanwhatever voltage is currently at the source terminal. However, it is highly recommended not to do so as will become apparent throughout the discussion.

68

a).

b).
MOSFETR
VDD

VGS

MOSFET
VGS

VDD

R

c).
MOSFET
VG

VDD
R

aaa-026549

Figure 7 | a) Low side MOSFET b) High side MOSFET driven Gate to Source c) High side MOSFET
driven gate to ground

In Figure 7c, when a VG is applied, first assumptions may be that the voltage acrossthe load resistor will be close to VDD due to the low on-state resistance of a
MOSFET compared to the load. However, the VGS of the MOSFET is VGS = VG – VS,
where VS is the voltage developed across the load in this example. This means thatas VG rises so, will VS thus reducing the overall VGS across the MOSFET. The result isa MOSFET that is not fully enhanced and therefore taking a significant portion of
the VDD across its drain and source terminals. The MOSFET is now dissipating aconsiderable amount of power and operating in linear mode, which may havethermal runaway implications. In addition, the load may not behave as expected
due to it no longer having the full VDD across it.

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

In Figure 7a the MOSFET source and gate-driver supply are both grounded andtherefore provides a true VGS supply. Likewise in Figure 7b, now the gate-drive isable to float above the source voltage and apply the full drive to the gate-source
terminals of the MOSFET. Both the circuits in Figure 7a and 7b the MOSFETs areeasily turned fully on and can achieve the low RDS(on). This allows the full VDD to be
applied across the load and ensures correct operation of the application.

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5.1.2 Half, Full and Three-Phase Bridges
Other application examples include half bridges and by extension full bridges andthree phase bridges which can be found in DC-to-DC convertors and motor drives.
Here, in any individual leg two MOSFETs are placed in series. When the MOSFETs
are off the resistance across the drain-source terminals is not infinite and leakage
paths exist. For simplification, these can be treated as resistances across thedrain-source terminals. Referencing Figure 8 it can be seen that the VGS on the highside (HS) MOSFET can be -VDD/2. The time the MOSFET VGS is at VDD/2 V may besignificant due to this occurring whenever the application is off and connected tothe supply voltage. This will need considering in the mission profile and the
MOSFET will need to be specified to meet this requirement.

VDD
HS MOSFET VGS:HS MOSFET

VG

VDD/2

VGS = VG - VS
VGS = 0 - VDD/2
VGS = - VDD/2

LS MOSFET

VGS

aaa-026550

Figure 8 | Half bridge MOSFET configuration showing leakage paths and potential issues if driving
High Side MOSFET with respect to ground

When the MOSFETs are driven with respect to ground the required gate-drive
needs to be at least the required VGS for the MOSFET summed with the maximum
voltage at the MOSFET’s source terminal. For example, an automotive application
typically uses a lead acid battery. Potentially the application could see 18 V fromthe battery and therefore the same voltage may appear on the source terminal of
the high side MOSFET. This means to drive a typical SL MOSFET a gate voltage of 28
V (18 V + 10 V) with respect to ground is required to ensure the MOSFET is fullyenhanced. Now the gate-drive is fixed at 28 V. However, the battery voltage may be
as low as 6 V. This means the VGS will be 22 V (28 V - 6 V). For a SL MOSFET this isoutside its safe operating limits and has the potential to reduce its reliability in theapplication.
Within half bridges the extremes of the VGS have been considered but the source
voltage of the high side MOSFET does not depend solely upon the battery-voltage:70

load conditions which are variable with temperature and supply-voltage result in
not knowing accurately what the source-voltage will be during considerable periodsof operation of the application. This makes it very difficult to assess the VGS
requirement of the MOSFET when driving them with respect to ground. It ispreferred and advisable to have the gate- driver referenced to the source of the
MOSFET and not ground. Now, when the source voltage changes the gate-drive will
be referencing this and the VGS will be controlled. However, even when driving the
MOSFET with respect to source there are still times when the VGS is not fully
dictated by the gate-driver. An assessment of transient analysis is required next.
All the applications discussed and their VGS behaviour have been assessed understeady-state conditions, the next few paragraphs take into consideration the
dynamic behaviour whilst the MOSFET is switching or commutating currents. In theexample figures already shown the gate-drivers have been idealised as perfectvoltage sources. However, all gate-drivers will have output impedances and drive
current limitations adding to the complexity of transient behaviour for theapplication. In addition to this the PCB layout will be a factor, with track
impedances contributing to the application behaviour. It is beyond the scope of thisapplication note to discuss best layout practices beyond noting that gate-drivers
should be placed as close as possible to the MOSFET it is driving. Some of theseissues have been addressed in application note AN11599 (chapter 3 of this book) Using power MOSFETs in parallel.
The transient operation of the half bridge is now reviewed, here solely the casewhere the MOSFET is driven gate to source is considered. The load is inductive,
representing half bridges (DC to DC convertors for example) and three-phasebridges (Brushless DC motors for example). The full operation of these applications
is not considered but in each, during the switching of the MOSFETs, theconsiderations to the gate-drive are similar.

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

Before describing the application behaviour, it is important to note that all
MOSFETs have parasitic capacitances that couple each of the terminals, gate, drain
and source (Figure 9). They are CGS, CGD and CDS and exist as part of the MOSFET
structure, playing a critical role in the transient behaviour of MOSFETs andtherefore the application.

CGD
MOSFET
CDS
CGS
aaa-026551

Figure 9 | Internal parasitic MOSFET capacitances71Nexperia | A Power Design Engineer's Guide

In half-bridges during an example transient event, one MOSFET is turned off andthe second is turned on, this causes the switch node voltage to rise and fall. The
switch node is where the source of the high side and the drain of the low side
MOSFETs are connected. The dV/dt on the switch node can couple through the CGD
of a MOSFET and charge or discharge that MOSFET’s CGS, appearing as VGS.
Whether the VGS is positive or negative depends on the sign of the dV/dt. It is thefunction of the driver to compensate for this change in VGS by charging/discharging
the VGS to the set level. How effective the gate-driver is at this depends on a coupleof factors, the rate of the dV/dt and how well the driver is coupled to the gate andsource (how much impedance is between them). Again, being aware that the total
impedance is a sum combination of package, resistance and inductances, trace
parasitics and intentionally placed components (which have even more parasitics).
This behaviour is shown in Figure 9. The consequences of this gate-bounce arebiasing of the gate for a cumulative time which needs to be considered in any
mission profile and worse, potentially causing cross conduction where both
MOSFETs are on simultaneously.

i=C

dV
dt

18
16

CGD
MOSFET
CDS
VGS

RG

Voltage (V)Designing in MOSFETs for safe and reliable gate-drive operation

2

CGS
VGS

L

MOSFET

14

VGS “bounces
back upduring the dV/dt

dV/dt across the highside FET causes acapacitive current tocharge CGD

12
10
8

HS VGS (V)6

LS VGS (V)4

Switch Node (V)20Some of this charge is sunk by the driver but some will
charge CGS and VGS will increase. Both MOSFETs canpotentially be on at the same time causing a shortcircuit and current to shoot through the MOSFETs

0

2

4

6

Time (µs)

8aaa-026552

Figure 10 | Dynamic behaviour of a MOSFET half bridge

5.1.3 High side MOSFETs with Inductive loads
Other applications where VGS behaviour may not be as first expected, are whendriving inductive-loads. Take again the examples in Figure 7 but now replace therestive-load with an inductive one. The difference in driving methodology, that iswith respect to ground or to the source, could result in either the MOSFET
operating in linear mode or entering avalanche respectively. Consider now thesituation where the gate-drive is referenced to ground. Also assume there issufficient drive on the gate to compensate for an increase in source voltage, thus
allowing the MOSFET to be fully enhanced.
When the MOSFET is turned on current will increase through the inductor, and72

energy will be stored in its magnetic field. Then the gate-driver is turned off, theinductor starts to demagnetise and produces a back-EMF in order to maintaincurrent flow. As one end of the inductor is grounded this back-EMF appears as anegative voltage on the source. The voltage will continue to decrease until the VGS
is great enough such that the MOSFET is partially turned on, maintaining theinductive current flow. Here the MOSFET is in linear mode, the VGS is somewhere
between off and fully on and the VDS is significant; remembering that the source is
now negative. See Figure 11 for an illustration.
Current (A)

15

5
0

VDD

MOSFET

0

20

40

60

Time (µs)

80

40

60

Time (µs)

8060Time (µs)

8060Time (µs)

8030
VG (V)

Gate

Inductor
Current

10Gate drive
turns off

20
10
0

0

20

SourceVG
Inductor
Current

L

VS (V)

20

Inductor forces source
voltage negative

10
0

-10

0

20

40

VGS (V)

12
Whilst the inductor
discharges the VGS is ~3 V

8
4
0

0

20

40

aaa-026553

Figure 11 | Circuit and waveforms showing the VGS behaviour whilst driving an inductive load and thegate of the MOSFET is driven gate to ground

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook

Taking the previous example of driving an inductive load but with the gate being
driven with respect to the source, the behaviour is similar until the MOSFET isturned off. Again the Vs will go negative until current flow is maintained, however
the source is tied firmly to the gate by the driver which is operating at 0 V. No
matter how negative the source goes (within operational limits) the MOSFET will
stay off. Instead, the Vs will continue to decrease and therefore the VDS willcontinue to rise. Ultimately the VDS will reach the breakdown voltage of the
MOSFET’s internal body diode and avalanche (Figure 12). As with linear mode thepower dissipation is considerable due to there being voltage and current present in
the MOSFET, but avalanche is a preferable method of absorbing the inductiveenergy. Compared to linear mode, avalanche operation is better at sharing thispower dissipation across the silicon die. Further information avalanche can befound in a MOSFET’s data sheet and in more detail in application note AN10273(chapter 4 of this book) - Power MOSFET single-shot and repetitive avalancheruggedness rating.

73

VDD

Inductor Current

5
0

0

5

10

15

Time (µs)

2015Time (µs)

2020Gate-drive turns off

100

Source

0

5

10

35
VS (V)

VGS

Inductor
Current

15
10

30
MOSFET

VG (V)

Gate

Current (A)Nexperia | A Power Design Engineer's Guide

L

The VDS will be VDD - VS.
This will be the breakdown
voltage of the MOSFET

The inductor forcessource voltage negative

15
-5
-25

0

5

10

15

Time (µs)

20

10

15

Time (µs)

20

12VGS (V)

Designing in MOSFETs for safe and reliable gate-drive operation

2

8

Whilst the inductor discharges
the VGS is held by the driver at 0 V
and the MOSFET avalanches

4
0

0

5

aaa-026554

Figure 12 | Circuit and waveforms showing the VGS behaviour whilst driving an inductive load and thegate of the MOSFET is driven Gate to Source

5.1.4 Motor Loads
One final application to be considered is driving motors with a single high side FET.
Again the designer is recommended to drive the MOSFET with respect to ground toensure full control of the MOSFET and application behaviour. Driving with respect
to ground and the source voltage will be near VDD due to motor back EMF; all theprevious considerations to driving respect to ground apply.

6 Application assessment summary
This chapter has summarised some common application behaviours and theirimpact on the MOSFET’s VGS. The application note has shown that driving a
MOSFET with respect to ground causes numerous issues and difficulties. These
include operating in linear mode and not knowing what the VGS may be at any point
in time. This makes it difficult to formulate a mission profile to select the correct
MOSFET. However, driving the gate with respect to source does not remove all of
the issues; there are transient events that can induce positive or negative voltages
on the VGS that are not fully controlled by the gate-driver. We are still short,however, from producing the mission profile to ensure the correct selection of a
MOSFET. The correct MOSFET should not be over-specified and should still operatereliably in the application for its intended life. We now need to create a mission
profile.

74

7 Creating a mission profile
An example mission profile is shown in Table 1, it is a distribution of VGS voltages at
a range of temperatures and for how long. However, before generating a profile,
several pieces of information need to be gathered. These include but are notlimited to:• How long the application is in operation– On and Off times– Possible fault conditions• The distribution of operating temperatures• The distribution of supply voltages
Along with an understanding of the application as outlined is the previous sections,this information can be used to create a mission profile. The mission profile can beassed against the TDDB data for a given MOSFET, thus determining the MOSFET’ssuitability for the application.

7.1 Example – Full H-bridge motor drive
An example automotive application is brushed Direct Current (DC) motors such as
windscreen wiper motors or seat control motors. For this worked example the
MOSFETs are driven with respect to their source terminals and not to ground.
The first part of the exercise is to gather information on how long the application
will be operating for. This information can then be translated into how long the
MOSFETs are on for and therefore how long they will have a VGS applied. In thisinstance it is common for the Original Equipment Manufacturer (OEM) to stipulate
the usage profile of the system and it may look like (Table 2).

2
Designing in MOSFETs for safe and reliable gate-drive operation

MOSFET & GaN FET Application Handbook

Table 2: Number of motor operations and the total time spent in operation
Operations

Total operation time at 0.5 s
per actuation (hours)

180,000

5

Other times when there may be a voltage present at the gate of the MOSFET iswhen the engine is off but the electronics are on. In the case of the motor, the twohigh side MOSFETs may be switched on to brake the motor. This time will add
cumulatively to the mission profile (Table 3).

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Designing in MOSFETs for safe and reliable gate-drive operation

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Nexperia | A Power Design Engineer's Guide

Table 3: Battery voltage whilst the engine is off and the time spent at thisvoltage whilst the electronic systems are connected to the battery
VGS (V)Time (hours)

12.5

6,000

Table 3 provides the first piece of information required to build our VGS profile. We
need to convert the information in Table 2 into the same. To do so, we need toknow the VGS the MOSFET will be driven at. The VGS may be set at 10 V by a power
supply or alternatively the MOSFET may be driven directly by the battery voltage.
The battery voltage will have its own distribution (Table 4). Using Table 2 and Table
4 we get Table 5.

Table 4: Battery voltage distribution of the life time of the application
Battery
voltage

Distribution

14.5

77%

16

15%

17

5%

18

3%

Table 5: VGS profile based on battery voltage distribution and application
operation time, including the VGS applied to the MOSFET whilst the engine
is off
VGS (V)Time (hours)

12.5

6000

14.5

19.25

16.0

3.75

17.0

1.25

18.0

0.75

20.0

0.00825

Table 5 is now a VGS profile, it shows the time spent at each VGS value. However,
it is missing the temperature profile to make it into a full mission profile to beassessed against the TDDB data. To generate a temperature profile the OEM andthe application designer need to work together. Remember that the MOSFET may
be operating a little hotter than its ambient (due to self-heating) so engineering
margin may need to be added. An example is shown in Table 6.

Table 6: Ambient and junction temperature distribution of the expected
life time of the application
The junction temperature is arbitrarily increased by 50 °C, this can be changed based onengineering judgment or calculation when better informed of the systems operationalparameters.
Tamb (ºC)

Tj (ºC)

Distribution

-40

10

6%

23

73

20%

40

90

65%

75

125

8%

80

130

1%

Table 5 contains the VGS values, and Table 6 has the temperature distribution.
These two together contain the information to make Table 7, the full VGS profile.
For example, for 6% of the time the MOSFET junction temperature is at 10 °C. By
multiplying the hours at each VGS (from Table 5) by 6% the Tj = 10 °C column inTable 7 is populated. It may be the case that the higher VGS voltages only occur atthe lower temperatures. The profile in Table 7 will need to be updated to reflect
this but can only be done so by the application designer working with the OEM.

Designing in MOSFETs for safe and reliable gate-drive operation

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MOSFET & GaN FET Application Handbook77Designing in MOSFETs for safe and reliable gate-drive operation

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Nexperia | A Power Design Engineer's Guide

Table 7. Full mission profile showing the time in hours for a given VGS
voltage and MOSFET junction temperature

VGS (V)Time at VGS and Temperature (hours)
Tj = 10 °C

Tj = °C

Tj = 90 °C

Tj = 125 °C

Tj = 130 °C

12.5

360

1200

3900

480

60

14.5

1.155

3.85

12.5125

1.54

0.1925

16

0.225

0.75

2.43750.3

0.0375

17

0.075

0.25

0.81250.1

0.0125

18

0.045

0.15

0.48750.06

0.0075The profile in Table 7 shows that for a time the MOSFET will see a VGS of 18 V atelevated temperatures. Using solely the datasheet ratings a logic-level MOSFET’s
VGS- max rating of 15 V would seem to exclude it from use in this application.
However, working with Nexperia to compare the profile to the logic level MOSFETs
TDDB data it can be shown that the MOSFET is more than capable of operating
reliably in the application. This is because the time spent at these conditions isshort.
Once the design engineer has generated a VGS mission profile they are encourage
to contact their local Nexperia representative to receive support. Working with
Nexperia, the profile can be assessed against the TDDB data and a suitable MOSFET
can then be suggested.

8 Summary
The gate-source lifetime and the gate-source turn-on voltage are highly dependent
on temperature so one needs to be careful in comparing datasheets from different
suppliers where the stated temperature conditions for the gate-source voltages
are different.
For devices that do not see the maximum gate-voltage at maximum temperature
throughout their intended life, Nexperia has a methodology to provide a better
assessment of the required gate-oxide rating for such an application. This would
often allow better performance in other parameters without compromisingreliability in the application.
This does require the so-called mission profile of the MOSFET in the application, as
described in this chapter.

78

Visit blog
Benefits of low Qrr
MOSFETs in motor
control applications

Benefits of low Qrr
MOSFETs in switching
applications

Visit blog

Visit blog

Automotive Trench 9Power MOSFETs designed
for performance andendurance

The right package for 12
48 V DC-to-DC conversion

Qrr overlooked andunderappreciated inefficiency battle

9 References

Designing in MOSFETs for safe and reliable gate-drive operation

2

MOSFET & GaN FET Application Handbook

1. Rating systems for electronic tubes and valves and analogous semiconductor
devices. IEC. IEC 60134:1961.
2. Failure Mechanism Based Stress Test Qualification For Discrete Semiconductors.
AEC. Vol. Rev D. AEC-Q101.
3. A methodology for projecting SiO2 thick gate oxide reliability on trench power
MOSFETs and its application on MOSFETs VGS rating. Efthymiou, E, Rutter, P andWhiteley, P. 2015, Microelectronics Reliability.

79Designing in MOSFETs for safe and reliable gate-drive operation

2

80

Nexperia | A Power Design Engineer's Guide

10 Tables
Table 1.
		
Table 2.
Table 3.
		
Table 4.
Table 5.
				
Table 6.
		
Table 7.
		

Example MOSFET mission profile, showing the time spent in .....................
hours at a particular VGS and MOSFET junction temperature................... 68
Number of motor operations and the total time spent in operation...... 75
Battery voltage whilst the engine is off and the time spent at this ...........
voltage whilst the electronic systems are connected to the battery...... 76
Battery voltage distribution of the life time of the application............... 76
VGS profile based on battery voltage distribution and application ............
operation time, including the VGS applied to the MOSFET whilst the .........
engine is off....................................................................................................... 76
Ambient and junction temperature distribution of the expected
life time of the application.............................................................................. 77
Full mission profile showing the time in hours for a given VGS
voltage and MOSFET junction temperature................................................ 78

11 Figures
Figure 1. VGS threshold values for an Nexperia automotive grade standard
level device ........................................................................................................61
Figure 2. VGS threshold values for an Nexperia automotive grade logic
level device.........................................................................................................62
Figure 3. Maximum VGS rating for a Nexperia automotive grade standard
level device.........................................................................................................63
Figure 4. Maximum VGS rating for a Nexperia automotive grade logic
level device.........................................................................................................63
Figure 5. Front page of a typical Nexperia automotive data sheet, showing the ......
MOSFET is automotive by highlighting that it is Q101 compliant............64
Figure 6. Cross-section of a power MOSFET at approximately 25,000 ........................
magnification showing where the MOSFET channel forms and the ...........
location of the gate-oxide. The gate-oxide is present both sides of
the trench and a channel forms next to both oxides. Current
flows vertically from Drain to Source through the MOSFET
channel................................................................................................................65
Figure 7. a) Low side MOSFETb) High side MOSFET driven Gate to Sourcec) High side MOSFET driven gate to ground.................................................69
Figure 8. Half bridge MOSFET configuration showing leakage paths andpotential issues if driving High Side MOSFET with respect toground.................................................................................................................70
Figure 9. Internal parasitic MOSFET capacitances........................................................71
Figure 10. Dynamic behaviour of a MOSFET half bridge..............................................72
Figure 11. Circuit and waveforms showing the VGS behaviour whilst driving an ..........
inductive load and the gate of the MOSFET is driven gate to ground.....73
Figure 12. Circuit and waveforms showing the VGS behaviour whilst driving an ..........
inductive load and the gate of the MOSFET is driven Gate to
Source..................................................................................................................74

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MOSFET & GaN FET Application Handbook

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Using power MOSFETs in parallel

3

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Nexperia | A Power Design Engineer's Guide

Chapter 3
Using power
MOSFETs in parallel
Application Note: AN11599

Using power MOSFETs in parallel

3

MOSFET & GaN FET Application Handbook

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Using power MOSFETs in parallel

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Nexperia | A Power Design Engineer's Guide

Increasing the capability of a MOSFET switch element by using several individual
MOSFETs connected in parallel can be useful. However, when designing switch
elements careful consideration of the circuit requirements and the MOSFET
characteristics and behavior must be applied. This chapter explores some of the
problems that can be experienced when several MOSFETs are operated in aparalleled group. It also suggests ways to avoid the problems and minimize theireffects.

1 Introduction
One much publicized benefit of power MOSFETs (compared to othersemiconductor devices) is that it is easy to parallel them to create a group withincreased capability. Although this feature is superficially true, there are several
potential problems that can catch out the unwary circuit designer.
A MOSFET consists of a group of paralleled ‘cells’ fabricated on the surface of asilicon die. All the cells are created at the same time under the same conditions.
When the MOSFET is fully enhanced and conducts channel current, thetemperatures of all the cells are very similar. As the cells are structurally andthermally closely matched, they share current and power well and the parametersof the MOSFET can be well-defined.
There is a range of values for the parameters of the MOSFET dies on a wafer. Thereis a wider range for MOSFET dies on different wafers in the same production batch.
The range is even wider for all batches even though the MOSFETs are the same
type. MOSFETs with parameter values outside the data sheet limits are rejected.
The MOSFETs in a paralleled group should all be the same type, but theirparameters could be anywhere within the data sheet range. Their die temperatures
are unlikely to be the same. Consequently their power sharing is not perfect.
This chapter contains guidelines on how to design a group of MOSFETs to get thebest performance from them. The design must accommodate MOSFET variations
within the data sheet limits. It must also allow for MOSFET parameter variations
over the range of electrical and environmental conditions. If all the MOSFETs in thegroup work within their safe maximum limits, the paralleled group of MOSFETs
operates reliably.
It is technically and commercially undesirable to have to select MOSFETs and it
should be unnecessary. The circuit should be designed to accommodate any
MOSFET within the worst case RDS(on) range.
The key data sheet limit which must not be exceeded is the maximum junction
temperature Tj(max) of 175 °C.

84

2 Static (DC) operation
This situation is the simplest condition where current flows through a group ofparalleled MOSFETs that are fully enhanced (switched ON). A proportion of thetotal current flows through each MOSFET in the group. At the initial point of turnon, the die temperatures of all the MOSFETs in the group are the same. The draincurrent ID flowing in each MOSFET is inversely proportional to its RDS(on) (thedrain-source voltage VDS across all the MOSFETs is the same).
The MOSFET with the lowest RDS(on) takes the highest proportion of the current anddissipates the most power (power dissipation P = VDS × ID).
All the MOSFETs heat up, but the MOSFET with the lowest RDS(on) heats up most(assuming the Rth(j-a) of all the MOSFETs is the same).

3
Using power MOSFETs in parallel

MOSFET & GaN FET Application Handbook

MOSFET RDS(on) has a positive temperature coefficient. RDS(on) increases as Tj
increases. The die temperatures and RDS(on) values of all MOSFETs in the group rise,
but the die temperature of the lowest RDS(on) MOSFET increases disproportionately.
The effect of this behavior is to redistribute the current towards the other higher
RDS(on) MOSFETs in the group.
Stable thermal equilibrium is reached after a period of operation. The lowest RDS(on)MOSFET is the hottest, but carries a lower proportion of the current than it did
initially.
The Positive Temperature Coefficient (PTC) of RDS(on) is a stabilizing influence that
promotes power sharing between the MOSFETs in the group. However, as statedearlier the most important criterion is that the maximum junction temperature of
any MOSFET in the group must not exceed 175 °C.
The cooling of each MOSFET in the group depends on its thermal resistance fromjunction to ambient. Die temperature influences the heat flow from adjacent
MOSFETs. Rather than considering the thermal resistance paths between the
MOSFET dies, the main influence is the temperature of the common heatsink. All
the MOSFET mounting bases are bonded electrically and thermally to this heatsink.
The lowest RDS(on) MOSFET could be located anywhere in the group. The thermal
resistance from mounting base to ambient of all the MOSFETs in the group should
be as similar as possible and as low as possible. Cooling is optimised andindependent of location.
This thermal resistance solely depends on the thermal characteristics and design of
the assembly [Printed-Circuit Board (PCB) or heatsink] to which the MOSFET isthermally bonded.

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Nexperia | A Power Design Engineer's Guide

Sometimes a value for Rth(j-a) is given in data sheets but this parameter is of very
limited value. It cannot be treated as a well-defined parameter because it also
depends on other external factors such as PCB construction, PCB orientation andair flow. The only guaranteed thermal parameter is the thermal resistance from the
MOSFET junction to mounting base Rth(j-mb).

2.1 Worked examples for static operation
The worked examples that follow are based on the BUK764R0-40E.

Table 1: Thermal characteristics
SymbolParameter

Rth(j-mb)

thermal resistancefrom junction tomounting baseRth(j-a)thermal resistancefrom junction toambient

Conditions

minimum footprint;
mounted on a PCB

Min

Max

Unit

-

-

0.82

K/W

-

50

-

K/W

A typical group of MOSFETs has a range of RDS(on) values. The distribution has apeak at around the typical data sheet RDS(on) value; none should have an RDS(on)higher than the data sheet maximum. The RDS(on) of about half of the samples is lessthan the typical value.
Minimum RDS(on) is not given in the data sheet, but a good estimate is
RDS(on)(min) ≈ RDS(on)(max) – 2(RDS(on)(max) – RDS(on)(typ) )

		

[1]

The RDS(on) value range means that a group of typical MOSFETs is very unlikely toshare power equally when they are operated in parallel.
The worst case would be when one of the MOSFETs has the minimum RDS(on) and all
the others have the maximum RDS(on).
Modeling of the electro-thermal system is complex because its electrical andthermal characteristics are mutually dependent. However, an electro-thermallyconvergent Excel model can be used to estimate the performance characteristics
of a paralleled group. As an example, in a worst case situation three BUK764R0-40E
MOSFETs are connected in parallel; two have maximum RDS(on) of 4 mΩ. The otherhas a lower than typical RDS(on) of 2.6 mΩ.
The MOSFET with the lowest RDS(on) value takes the highest proportion of thecurrent. It therefore has the highest power dissipation.

86

The target is to keep the junction temperature of the hottest MOSFET below
175 °C under worst case operating conditions.
These estimations are simplified illustrations. The thermal representation of the
MOSFET group is less complex than a real application. In a real application, there
are other factors such as neighboring components and orientation that would
influence cooling. However, they show the approximate behavior of the group withtwo different Rth(j-a) values.
The first scenario shows the system with the thermal resistance from junction toambient for each MOSFET (Rth(j-a) = 20 K/W; Tamb = 125 °C).

Using power MOSFETs in parallel

3

MOSFET & GaN FET Application Handbook

aaa-016409

Figure 1 | Electrical and thermal schematic diagram of the three paralleled BUK764R0-40E MOSFETs

Table 2 shows the maximum safe limits of VDS, ID, P and Tj for all the MOSFETs atthermal equilibrium. At this point, the junction temperature of the hottest MOSFET
is almost 175 °C.

MOSFET

VDS[V]

Rth(j-a)[K/W]

RDS(on)(25 °C)[mΩ]

InitialRDS(on)(125 °C)[mΩ]

Initialcurrent
ID
[A]

M1

0.11

20

2.6

4.16

42.31

M2

0.11

20

4

6.4

27.50

0.11

20

4

6.4

27.50

Final
RDS(on)[mΩ]

Finalcurrent
ID[A]

M3
Initial
power
P
[W]

Initial
power
share
[%]

Final
power
P
[W]

Final
Tj
[°C]

Final
power
share
[%]

4.65

43.5

4.92

22.36

2.46

174

42.3

3.03

28.3

7.21

15.26

1.68

159

28.9

3.03

28.3

7.21

15.26

1.68

159

28.9

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Nexperia | A Power Design Engineer's Guide

Total initial power P(M1 + M2 + M3) = 10.70 W
Total final power P(M1 + M2 + M3) = 5.82 W
The second scenario relates to the same electrical system, but with ideal thermal
characteristics. The thermal resistance Rth(j-a) of each MOSFET is 0.82 K/W. Thissituation corresponds to the ideal but unrealistic situation where each MOSFET isperfectly thermally bonded to an infinite heatsink (a heatsink with zero thermal
resistance).

aaa-016503

Figure 2 | Electrical and thermal schematic diagram of the three paralleled BUK764R0-40E MOSFETs
with ideal thermal conditions

Table 3 shows the values of VDS, ID, P and Tj for all the MOSFETs at thermal
equilibrium. At this point, the junction temperature of the hottest MOSFET isalmost 175 °C.

Table 3: MOSFET maximum conditions for Rth(j-a) = 0.82 K/W
and Tamb = 125 °C

88

MOSFET

VDS[V]

Rth(j-a)[K/W]

RDS(on)(25 °C)[mΩ]

InitialRDS(on)(25 °C)[mΩ]

Initialcurrent
ID
[A]

M1

0.55

0.82

2.6

4.16

211.54

M2

0.55

0.82

4

6.4

137.50

M3

0.55

0.82

4

6.4

137.50Table 3: continued...
Initial
power
P
[W]

Initial
power
share
[%]

Final
RDS(on)[mΩ]

Finalcurrent
ID
[A]

Final
power
P
[W]

Final
Tj
[°C]

Final
power
share
[%]

116.3543.5

4.92

111.79

61.48

174

42.3

75.63

28.3

7.21

76.28

41.96

159

28.9

75.63

28.3

7.21

76.28

41.96

159

28.9

Total initial power P(M1 + M2 + M3) = 267.60 W
Total final power P(M1 + M2 + M3) = 145.39 W
Note: This scenario is ideal and unrealistic. It is included to illustrate the benefit ofreducing Rth(j-a) to maximize the usage of the MOSFET capabilities.

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3

MOSFET & GaN FET Application Handbook

In practice, Rth(j-a) is always greater than Rth(j-mb). The thermal bonding between the
MOSFET mounting base and the heatsink is never perfect and an infinite heatsink
does not exist in the real world.
In this case, the drain current (ID) of the MOSFETs becomes the limiting factor (thedata sheet maximum ID is 75 A).
To optimize MOSFET utilization, the MOSFETs must be able to dissipate as much
power as possible. At the same time, the junction temperature of the hottest
MOSFET must remain below the maximum safe temperature of 175 °C.
The following conclusions can be drawn from Table 2 and Table 3:
1. It is beneficial to reduce Rth(j-a) as much as possible to optimize the MOSFET die
cooling.
2. It is beneficial to reduce the maximum ambient temperature to increase theavailable thermal 'headroom'.
3. As a result of Table 2 and Table 3, it is clear that junction temperature difference
between the MOSFETs depends only on their RDS(on) values (assuming the Rth(j-a) for
each MOSFET is the same).
4. When the paralleled (fully enhanced) MOSFETs heat up during use, their power
distribution changes such that the cooler MOSFETs take a greater proportion of thepower. This effect is due to the PTC of RDS(on) and it acts to promote thermal
stability.

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Nexperia | A Power Design Engineer's Guide

3 MOSFET mounting for good thermal
performance and power sharing
To get the most from the MOSFET group, the individual MOSFET should be
mounted in a way that causes their mounting base temperatures to be as similar as
possible and also as low as possible.
To realize this goal, the thermal resistance between each MOSFET (mounting base)and the mounting bases of all the other MOSFETs in the group should be matched
and minimized. They should be mounted symmetrically and as close together as
possible on a thermally conductive surface.
Heat flow can be considered to be analogous to electric current flow; so thethermal bonding points of the MOSFETs (usually the drain tabs) should be on athermal ‘ring main’. The low thermal resistance path allows heat to flow easily
between the MOSFETs. When heat flows easily between all the MOSFETs in thegroup, their mounting base temperatures track together closely.
Note: This arrangement does not promote equal current sharing, but promotes
better die temperature matching. The temperatures of all the MOSFETs in thegroup can rise more before the temperature of the hottest MOSFET reaches
175 °C. Hence the power dissipation capability of the group is maximized.
There are practical limits to the physical extent of the thermal ring main; ideally
each MOSFET should be next to all its neighbors. This condition limits the group to
two or three MOSFETs.

This layout suits standard packaged MOSFETs (e.g. D2PAK, LFPAK and TO-220). The dark shaded area corresponds to thedrain tab of the device.
aaa-016511

Figure 3 | A layout for six paralleled MOSFETs to promote good power sharing

90aaa-016512

a. Two paralleled MOSFETs
where both devices arelocated directly opposite
each other on the two facesof a PCB

aaa-016513

b. Two paralleled MOSFETs
where the devices arelocated next to each other on
the same face of a PCB

aaa-016514

c. Three paralleled MOSFETs
where the devices arelocated in a symmetrical loop
other on one face of a PCB

Figure 4 | Good layout arrangements for two paralleled MOSFETs and three paralleled MOSFETs

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3

MOSFET & GaN FET Application Handbook

A good way to parallel a pair of MOSFETs is to locate them on opposite faces of a
PCB forming a PCB ‘sandwich’ as in Figure 4a. Thermal vias between the copper‘land’ areas on the PCB reduce the electrical and thermal resistances between theirmounting bases.
To parallel a pair of MOSFETs on the same face of a PCB, they can be mounted next
to each other as in Figure 4b.
A good way to parallel three MOSFETs is in a ring as in Figure 4c. This arrangement
allows all the MOSFET sources in the group to be connected to a ‘star’ point. The
electrical and thermal paths between the MOSFET drains match due to theirsymmetrical connections to the drain loop.
Minimizing and matching the electrical impedances in the source paths is more
important than matching the electrical impedance in the drain paths. Thisdifference is because their gate drives are related to the sources. Good impedance
matching in both drain and source is more important in high frequency switchingcircuits.
If similar electrical and thermal matching can be achieved, larger paralleled groups
could be considered. Groups of more than four or five become unwieldy sogrouped sub groups should be used.

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aaa-016511

a. Six paralleled MOSFETs based on two sub groupsof three MOSFETs

aaa-016518

b. Nine paralleled MOSFETs based on three
sub groups of three MOSFETs

Figure 5 | Layouts using sub groups of three MOSFETs

In this group of 9 (3 groups of 3), there is a natural drain star point at the center ofthe group. There are separate source star points at the centers of each sub group.
The source star points can be connected on a layer of a multi-layer PCB.
There is a compromise between optimizing layout for power sharing andmaximizing PCB usage.
Electrically and thermally optimised layouts always use more PCB area than theminimum possible, but utilization of MOSFET capability should be better. Areas of
PCB that are unoccupied by MOSFETs or gate drive components are useful forthermal interfaces with heat sink or cooling air.

4 Power sharing in dynamic operation [pulse
and Pulse Width Modulation (PWM) circuits]
Many MOSFET circuits are designed to operate in systems where they are switchedrepetitively (such as DC-to-DC converters). Paralleled MOSFETs can be used as theswitching elements in the system, but in addition to the guidelines set out for
optimal steady state power sharing. Some additional points must be considered sothat the MOSFETs share current during the switching transitions.
Good circuit and layout design is important. It influences the proportion of current
carried by each MOSFET in the group during and after the switching event.

92

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aaa-016505

Figure 6 | Schematic diagram showing the stray source inductances and source resistances

aaa-016507

a. MOSFET drain current distribution immediately
after turn on. The time base scale is
100 ns/division.

aaa-016508

b. The same MOSFET drain current distribution as in
Figure 7a showing the changes as time progress.
The time base scale is 100 µs/division.

Figure 7 | Oscilloscope plots showing the MOSFET drain currents (black, red and blue) and drain
source voltage (yellow) after turn on

Figure 7a and Figure 7b show how the current distribution in three paralleled
MOSFETs initially depends on the source inductances in the MOSFET current paths,
namely L1, L2 and L3. As time progresses the resistances in the MOSFET current
paths, namely RDS(on)+ R1, RDS(on) + R2 and RDS(on)+ R3 determine the current
distribution.
The same principles hold about impedance matching of the current paths to the
MOSFETs in the group. In this case, it is more important for the rates of change of93

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current in the MOSFETs to match. The source inductances are the key impedance as
they affect the gate-source voltages (gate drives) of the MOSFETs in the group.
This effect dominates more in high frequency and short duty cycle applications (e.g.
switched-mode power supplies). It may be insignificant in lower frequency
applications such as motor drives.

5 Partially enhanced (linear mode) power
sharing
If a group of MOSFETs must operate in linear (partially enhanced) mode, great
caution is needed. MOSFETs simply paralleled together as they are for fullyenhanced conduction are very unlikely to share power or current well.
This behavior is due to the Negative Temperature Coefficient (NTC) of gate
threshold voltage VGS(th). As the group of MOSFETs starts to enhance, the MOSFET
with the lowest VGS(th) starts to conduct channel current first. It dissipates more
power than the others and heat up more. Its VGS(th) decreases even further whichcauses it to enhance further.
This unbalanced heating causes the hottest MOSFETs to take a greater proportionof the power (and get even hotter). This process is unsustainable and can result in
MOSFET failure if the power is not limited. Great care is needed when designingparalleled power MOSFET circuits that operate in the partially enhanced (linearmode) condition.
If all the MOSFETs in the group operate within their Safe Operating Area (SOA),
they work reliably. The SOA must be adjusted for the worst case mounting basetemperature that occurs in the application. Remember that the data sheet SOAgraph applies only if the MOSFET mounting base temperature is 25 °C or less.
Adding external source resistors (R1 to R4 in the schematic; see Figure 8) provides
the negative feedback needed for stable operation. The gate-source voltage
applied to VGS (M1) = VG − ID (M1) × R1.
If the MOSFETs must also operate in fully enhanced mode (e.g. in ‘Hot Swap’ or‘Soft Start’ applications), the inclusion of these resistors is an efficiency
disadvantage.
As the MOSFET channel current increases, its gate drive voltage reduces.
As the MOSFETs are operating in partial enhancement (where MOSFET RDS(on) is notimportant), there is no adverse effect caused by including these resistors. If the
MOSFETs must operate in both modes (as with active clamping after fully enhanced
conduction), the inclusion of source resistors does have a negative impact.

94

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Figure 8 | Schematic diagram of a group of four paralleled MOSFETs intended to operate in linear
mode

MOSFETs developed specifically for linear mode operation are available from
Nexperia. They can simplify the design of paralleled groups of MOSFETs intended
for linear mode operation.

6 Gate drive considerations
It is preferable to fit low value gate resistors between the gate driver and the gateof each MOSFET in the group.
Their main function is to decouple the MOSFET gates from each other so they all
receive similar gate drive signals. Without these resistors, at turn on the Miller
plateau of the MOSFET with the lowest threshold voltage would act to clamp thegate-source voltages of the other MOSFETs in the group. This clamping effecttends to inhibit and delay the turning on of the other MOSFETs in the group. At
turn off a similar process occurs.
Without these resistors, the MOSFET with the lowest threshold voltage would
switch on first and switch off last. The consequences of this effect may be insignificant in low frequency high duty cycle applications. In higher frequency PWM
applications, it could cause a significant power imbalance between the MOSFETs.
Positive feedback also occurs in this case which increases the imbalance and could
ultimately cause MOSFET failure.
Gate resistors also help to damp out oscillatory transients on VGS. They also swamp
any effects caused by variations in the internal gate resistance RG(int) of the
MOSFETs.

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6.1 Should individual gate drivers be used foreach MOSFET in the group?
Using individual gate drivers for each MOSFET in the group is usually unnecessary.
They may be necessary in applications where fast switching of a large group oflarge die MOSFETs is needed. Here the MOSFETs should be arranged in smaller subgroups, each sub group driven by an individual gate driver. Care should be taken tobalance the circuit so the propagation delays of all the gate drivers are similar. This
matching ensures that the switching of all the MOSFETs in the group is synchronized. Usually it is sufficient to drive the gate of each MOSFET in the group fromthe same gate driver. However, it is important to have a gate resistor between thegate driver output and the gate of each MOSFET as mentioned earlier.

7 MOSFET packaging considerations forparalleled groups
Conventionally packaged surface-mounted MOSFETs (DPAK and D2PAK) are themost widely available types so they are considered first for paralleled groups.
However, KGD and LFPAK (power SO8) MOSFETs could offer better solutions.

7.1 Bare die (KGD) MOSFETs
These MOSFETs offer the densest and most flexible options for paralleled groups;
they are designed to suit a specific application. The die aspect ratio and gate padlocation can be designed specifically to suit the application. More source wirebonds can be fitted to the die than can be fitted in a conventionally packaged
MOSFET, so overall RDS(on) can be reduced. Maximum drain current can be increased
to achieve better performance from a paralleled group of MOSFETs. Special
manufacturing facilities are required for KGD assembly.

7.2 LFPAK MOSFETs
The power SO8 (LFPAK) MOSFETs offer the opportunity to manufacture theparalleled MOSFET circuit conventionally. Higher component density and power
capability are possible (approaching that of KGDs). The connections to the source
and gate are made using copper clips which give better electrical and thermal
performance than aluminum wire bonds in conventional packages.

96

8 Inductive energy dissipation in paralleled
MOSFETs
8.1 Avalanching - low side MOSFET groupdriving a high side inductive load
If the group of paralleled MOSFETs is driving an inductive load, energy stored in thisload must be safely dissipated when the current is switched off. A good way tomanage this energy is to connect a ‘freewheel diode’ across the load; see Figure 9.
Current flowing in the MOSFET channel diverts into the diode when the MOSFETs
switch off and the energy is dissipated in the circuit resistances. However, it is notalways possible and energy must then be dissipated safely in the MOSFETs.

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MOSFET & GaN FET Application Handbook

If the battery polarity (Vsup) is reversed, the low impedance path through thefreewheel diode and the body diode can carry large damaging currents. Freewheel
diodes are often not used for this reason.
When the group of MOSFETs is switched off, the back e.m.f. from the inductive load
may be high enough to cause the drain-source voltage across the group of
MOSFETs to exceed the drain-source breakdown voltage V(BR)DSS of one of the
MOSFETs. It likely that MOSFETs in the group have a range of V(BR)DSS values (eventhough they are the same type). The current then flows through the body diode of
the MOSFET with the lowest V(BR)DSS in reverse (avalanche) conduction. Thiscondition causes high-power dissipation and temperature rise in the MOSFET die(P = ID × V(BR)DSS ). If the maximum 175 °C junction temperature is exceeded, thethermal stress on the die could degrade or destroy the MOSFET.
In the worst case, all the current which was flowing through the group of MOSFETs
could be diverted into the body diode of one MOSFET in the group. If this scenario
is possible, it is vital that a single MOSFET in the group can safely handle the total
avalanche current under worst case thermal conditions. V(BR)DSS has a positive
temperature coefficient which tends to redistribute the current towards other
MOSFETs with higher V(BR)DSS values.

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Using power MOSFETs in parallel3

aaa-016520

Figure 9 | Low side MOSFET group driving a high side inductive load

8.2 Active clamping - high side MOSFET groupdriving a low side inductive load
This configuration (see Figure 10) is often used in automotive applications. This
topology is useful because the vehicle chassis can be used as the negative supply
return path to the battery.

aaa-016521

Figure 10 | High side MOSFET group driving a low side inductive load

In this circuit, the difference between the threshold voltages rather than V(BR)DSS
spread determines where the load current flows. The MOSFET with the lowest
threshold voltage conducts the greatest proportion of the current. The

98drain-source voltage across the MOSFET group is Vsup + VGS and the power
dissipation of the group is (Vsup + VGS) × ID. As with the avalanche case, all thecurrent (and hence all the power dissipation) could be diverted into a single
MOSFET in the group.
This situation is worse than the avalanche case because MOSFET threshold voltage
has a negative temperature coefficient. This characteristic tends to direct thecurrent flow initially to the hottest MOSFET. This MOSFET then gets even hotter sothat it retains the current.

9 Summary
1. It is better to use a single large MOSFET rather than a group of smaller MOSFETs.

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MOSFET & GaN FET Application Handbook

2. The power capability of a group of n MOSFETs never achieves n times the power
capability of a single MOSFET.
3. If it is necessary to use paralleled MOSFETs, use the lowest number possible as
the basic group size (3 maximum).
4. If a larger number is needed, use a group of basic groups i.e. 4 = 2 groups of 2;
6 = 2 groups of 3.
5. The circuit layout is a very important factor determining how well a group ofparalleled MOSFETs share power dissipation, particularly in higher frequency
repetitive switching circuits.
6. Consider LFPAKs (for repetitive switching applications) because of their small
size, good thermal performance and low package impedances.
7. Special care is needed when designing groups of MOSFETs that could operate inavalanche or active clamping mode.

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10 Tables
Table 1.
Table 2.
Table 3.

Thermal characteristics ....................................................................................86
MOSFET maximum conditions forRth(j-a) = 20 K/W and Tamb = 125 °C...................................................................87
MOSFET maximum conditions forRth(j-a) = 0.82 K/W and Tamb = 125 °C........................................................ 88 - 89

11 Figures
Figure 1. Electrical and thermal schematic diagram of the three paralleled ..............
BUK764R0-40E MOSFETs................................................................................ 87
Figure 2. Electrical and thermal schematic diagram of the three paralleled ..............
BUK764R0-40E MOSFETs with ideal thermal conditions........................... 88
Figure 3. A layout for six paralleled MOSFETs to promote good power
sharing................................................................................................................ 90
Figure 4. Good layout arrangements for two paralleled MOSFETs and three ............
paralleled MOSFETs ......................................................................................... 91
Figure 5. Layouts using sub groups of three MOSFETs ............................................. 92
Figure 6. Schematic diagram showing the stray source inductances andsource resistances ........................................................................................... 93
Figure 7. Oscilloscope plots showing the MOSFET drain currents(black, red and blue) and drain source voltage (yellow) afterturn on................................................................................................................ 93
Figure 8. Schematic diagram of a group of four paralleled MOSFETs intended
to operate in linear mode ............................................................................... 95
Figure 9. Low side MOSFET group driving a high side inductive load ..................... 98
Figure 10. High side MOSFET group driving a low side inductive load...................... 98

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Chapter 4Power MOSFET single-shotand repetitive avalanceruggedness rating
Application Note: AN10273

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1 Introduction
Power MOSFETs are normally measured based on single-shot Unclamped Inductive
Switching (UIS) avalanche energy. This chapter describes in detail, the avalancheruggedness performance, fundamentals of UIS operation and appropriatequantification method for the safe operating condition.
Electronic applications have progressed significantly in recent years and haveinevitably increased the demand for an intrinsically rugged power MOSFET. Device
ruggedness defines the capacity of a device to sustain an avalanche current during
an unclamped inductive load switching event. The avalanche ruggedness
performance of a power MOSFET is normally measured as a single- shot Unclamped
Inductive Switching (UIS) avalanche energy or EDS(AL)S. It provides an easy and quick
method of quantifying the robustness of a MOSFET in avalanche mode. However, it
does not necessarily reflect the true device avalanche capability (see Ref. 1, Ref. 2
and Ref. 3) in an application.
This chapter explains the fundamentals of UIS operation. It reviews the appropriate
method of quantifying the safe operating condition for a power MOSFET, subjected
to UIS operating condition. The chapter also covers the discussions on repetitive
avalanche ruggedness capability and how this operation can be quantified tooperate safely.

2 Single-shot and repetitive avalanche
definitions
Single-shot avalanche events are avalanche events that occur due to a fault
condition in the application such as electrical overstress. The application does nothave an avalanche designed into its operation.
However, repetitive avalanche refers to the applications where avalanche is anintended operation mode of the MOSFET. Here, avalanche is a designed function
and has a limited number of operations to ensure reliability of the MOSFET over its
life. The number of allowed events is a function of the avalanche energy and can be
determined from the data sheet charts found on repetitive avalanche specific parts– ending in suffix R e.g. BUK9K51-60R.
Any customer wishing to operate outside the current avalanche ratings may beconsidered on an application basis. Contact your local sales team for more
information.

104

3 Understanding power MOSFET single-shot
avalanche events
The researchers and the industry have established single-shot avalanche capabilityof a device (see Ref. 1, Ref. 2 and Ref. 3). The test is carried out on a simple
unclamped inductive load switching circuit, as shown in Figure 1.

1
L

drain
2
gate
gate voltageRsourceVDD
001aaj764

Figure 1 | Unclamped inductive load test circuit for MOSFET ruggedness evaluation

3.1 Single-shot UIS operation
A voltage pulse is applied to the gate to turn on the MOSFET, as shown in Figure 2.
It allows the load current to ramp up according to the inductor value (L) and thedrain supply voltage (VDD). The phenomenon is shown in Figure 3 and Figure 4. At
the end of the gate pulse, the MOSFET is turned off. The current in the inductor
continues to flow, causing the voltage across the MOSFET to rise sharply. This
overvoltage is clamped at breakdown voltage (VBR) until the load current reaches
zero, as illustrated in Figure 3. Typically, VBR is:

Power MOSFET single-shot and repetitive avalanche ruggedness rating

4

MOSFET & GaN FET Application Handbook								 [1]The peak load current passing through the MOSFET before turn off is the nonrepetitive drain- source avalanche current (IDS(AL)S) of the UIS event. IDS(AL)S is
illustrated in Figure 4. The following expression is used to determine the rate at
which the avalanche current decays, which is dependent on the inductor value:
				

[2]

			
The peak drain-source avalanche power (PDS(AL)M) dissipated in the MOSFET is105

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shown in Figure 5. It is a product of the breakdown voltage (VBR) and the nonrepetitive drain-source avalanche current (IDS(AL)S); see Figure 3 and Figure 4. The
avalanche energy dissipated is the area under the PAV waveform and is estimatedfrom the following expression:							 [3]or			Another crucial parameter involved in a MOSFET avalanche event is the junction
temperature. After the avalanche event (τ) has begun, the following expression isused to determine the transient junction temperature variation during device
avalanche at a given time:
				

[5]

where Zth is the power MOSFET transient thermal impedance. Alternatively, thefollowing expression approximates the maximum ΔTj:
				

[6]

Assuming that Tj(max) occurs at tAL/2, Zth(tAL) ⁄ 2 is the transient thermal impedance
measured at half the avalanche period tAL. Note, the Zth value used for theavalanche calculation is more conservative than the one published in data sheets
due to the nature of avalanche.
Therefore, the maximum junction temperature resulting from the avalanche event
is:
				
[7]
where Tj refers to the junction temperature prior to turn off.

106

[4]

3.1.1 Single-shot UIS waveforms

VDS

VGS

VBR
V(BR)DSS

ON

VDD

OFF

VDD

time
001aaj765

tAL

Figure 2 | Gate-source voltage,VGS			

Figure 3 | Drain-source voltage, VDS

ID

PIDS(AL)SPDS(AL)M

area in PDS(AL) triangle= avalanche energy(EDS(AL)S) dissipatedtime
001aaj767

tAL

time

001aaj766

time
tAL

Figure 4 | Drain current, ID			

001aaj768

Figure 5 | Peak drain-source avalanche					power, PDS(AL)M

Tj
Tj(max)Tj(max)
Tj

tAL

time
001aaj769

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MOSFET & GaN FET Application Handbook

Figure 6 | Transient junction temperature profile of MOSFET during an avalanche event

3.2 Single-shot avalanche ruggedness rating
The failure mechanism for a single-shot avalanche event in a power MOSFET is dueto the junction temperature exceeding the maximum temperature rating. In such acase, catastrophic damage occurs to the MOSFET. If the transient temperature
resulting from an avalanche event, as shown in Figure 6, rises beyond a
recommended rated value, the device risks being degraded. The recommended
rated value is de-rated from the maximum temperature for optimum reliability.
Blackburn (see Ref. 2) has discussed a general guideline in detail, on the107Power MOSFET single-shot and repetitive avalanche ruggedness rating

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appropriate method of quantifying the single-shot avalanche capability of a device.
It takes the avalanche current and initial junction temperature into consideration.
The maximum allowed avalanche current as a function of avalanche time defines
the safe operation for a device single-shot UIS event. The maximum allowed
avalanche current is set so that a safe maximum junction temperature, Tj(max) of
175 °C, is never exceeded. Using Equation 7, Figure 7 is plotted.
102
IAL
(A)

10

1

10-1
10-3

10-2

10-1

1
tAL (ms)10

aaa-031581

Figure 7 | BUK9K51-60R avalanche rating; avalanche current as a function of avalanche time

Figure 7 shows the SOA curve of a device single-shot avalanche capability. The
25 °C junction temperature curve shows the maximum allowable IAL for a given tALat an initial Tj of 25 °C. This maximum IAL results to a maximum allowable junction
temperature Tj(max) of 175 °C, which means a ΔTj(max) of 150 °C. The area under the
SOA curve is the Safe Operating Area (SOA).
The maximum junction temperature resulting in catastrophic device avalanche
failure is approximately 380 °C, which is in excess of the rated Tj(max) of 175 °C.
However, operating beyond the rated Tj(max) may induce long-term detrimental
effects to the power MOSFET and is not recommended.

4 Understanding power MOSFET repetitive
avalanche events
Repetitive avalanche refers to an operation involving repeated single-shot
avalanche events, as discussed earlier. Until recently, most manufacturers haveavoided the issues pertaining to the power MOSFET repetitive avalanche capability.

108

It is primarily due to the complexity in such operations and the difficulties in
identifying the underlying physical degradation process in the device.
Due to the traumatic nature of the avalanche event, a repetitive avalanche
operation can be hazardous for a MOSFET. It is hazardous even when the individual
avalanche events are below the single-shot UIS rating. This type of operation
involves additional parameters such as frequency, duty cycle, and thermal
resistances Rth(j-a) and Rth(j-mb) of the system during the avalanche event. However, it
is possible to de-rate the single-shot rating to define a repetitive avalanche SOA.

4.1 Repetitive UIS operation
The repetitive UIS test circuit is shown in Figure 1. The gate is fed with a train ofvoltage pulses at a frequency (f) and for a duty cycle as shown in Figure 8. The
resulting breakdown voltage (VBR) and drain current (ID) passing through the load
are the same as for a single-shot UIS. However, the peak ID is now denoted as
repetitive drain-source avalanche current (IDS(AL)R), as shown in Figure 9.
The repetitive drain-source avalanche power (PDS(AL)R) resulting from the repetitive
UIS operation is shown in Figure 10. For finding the value of PDS(AL)R, it is necessary
to first calculate EDS(AL)S for a single avalanche event using Equation 3. This
resultant value of EDS(AL)S is substituted in the following expression, to calculate thevalue of PDS(AL)R:PDS(AL)R = EDS(AL)S x f

			

[8]

4.1.1 Repetitive UIS waveforms

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VGS
VBR
V(BR)DSS
IDS(AL)R

VDD
time

time
001aaj771

tAL

001aaj772

Figure 8 | Gate pulse, VGS

Figure 9 | Drain-source voltage, VDS and repetitive		drain- source avalanche current, IDS(AL)R109

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Tj

P

PDS(AL)M
PDS(AL)R

Tj

Tj(init)Tmb

Tj(AV)

time

time

tAL

tAL

001aaj773

001aaj774

Figure 10 | Repetitive drain-source avalanche

Figure 11 | Transient junction temperature

power, PDS(AL)Rcomponents of MOSFET during repetitive
avalanche

4.2 Temperature components
The temperature rise from the repetitive avalanche mode in the power MOSFET isshown in Figure 11.
The temperature (Tj(init)) comprises the mounting base temperature (Tmb) and thetemperature rise resulting from any on-state temperature difference (ΔTon).
Tj(init) = Tmb + ΔTon

			

[9]

In addition, there is a steady-state average junction temperature variation (ΔTj)
resulting from the average repetitive avalanche power loss.
ΔTj = PDS(AL)R x Rth(j-a)

					

[10]

where Rth(j-a) is the thermal resistance from junction to ambient of the device in theapplication. The summation of Equation 9 and Equation 10 gives the averagejunction temperature, Tj(AV) of a power MOSFET in repetitive UIS operation.
Tj(AV) = Tj(init) + ΔTj

				 [11]5 Repetitive avalanche ruggedness rating
Our investigations show that there is more than one failure or wear-out mechanism
involved in repetitive avalanche. Temperature is not the only limiting factor to arepetitive avalanche operation. However, by limiting temperature and therepetitive drain-source avalanche current (IDS(AL)R), an operating environment isdefined such that the avalanche conditions do not activate device degradation. It
allows the power MOSFET to operate under repetitive UIS conditions safely.

110Figure 12 shows the repetitive avalanche SOA curve of BUK9K51-60R, where foreach avalanche event the Tj rise is limited to 30 K.

IAL
(A)

102

10

1

10-1
10-3

10-2

10-1

1
tAL (ms)10

aaa-031582

Figure 12 | Repetitive avalanche rating; avalanche current

The three conditions which must be satisfied for safe operation of a power
MOSFET under repetitive avalanche mode are:
1. I(AL) should not exceed the repetitive avalanche SOA curve
2. Tj should not exceed 175 °C
3. The number of cycles should not exceed the avalanche cycle limit chart
The number of allowable cycles can be determined through a second chart in theavalanche MOSFET’s data sheet, (see Figure 13).
1010
No. Events

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109

108

1

10
EDS(AL) Per Event (mJ)102

aaa-031584

Figure 13 | Repetitive avalanche rating; maximum number of avalanche events as a function ofavalanche energy111

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6 Conclusion
Power MOSFETs can sustain single-shot and repetitive avalanche events. Simple
design rules and SOA regions are provided.

7 Examples
The following examples examine cases of avalanche operation acceptance:

7.1 Single-shot avalanche case
•
•
•
•
•
•

Device: BUK9K51-60R; see Figure 7
L = 30 μH
IAL = 18 A
Rth(j-a) = 5 K/W
V(BR)DSS = 60 V
VDD = 0 V

7.1.1 Calculation steps
1. Using the above information, tAL can be determined using Equation 2, whichin this case is 7 μs. Transferring the IAL and tAL conditions onto Figure 12, theoperating point is under the SOA curve. It suggests that the operating 		condition may be feasible.

7.2 Repetitive avalanche case
•
•
•
•
•
•
•
•

Device: BUK9K51-60R; see Figure 12 and Figure 13
L = 30 μH
I(AL) = 10 A
f = 3 kHzRth(j-a) = 5 K/W
To = 100 °C
V(BR)DSS = 60 V
VDD = 0 V

7.2.1. Calculation steps
1. From the above information, tAL can be determined using Equation 2, whichin this case is approximately 4 μs. Transferring the IAL and tAL conditions onto

112

2.
3.
4.
5.

6.

Figure 12, the operating point is under the boundary of the ‘Rep. Ava’ SOA
curve. It suggests that the operating condition is acceptable. Therefore, 		
condition 1 is satisfied.
Calculate the non-repetitive drain-source avalanche energy (EDS(AL)S) using
Equation 3 EDS(AL)S = 1.56 mJ).
Calculate the repetitive drain-source avalanche power (PDS(AL)R) using Equation
8 (PDS(AL)R = 4.68 W).
Calculate the average ΔTj rise from repetitive avalanche (ΔTj) using Equation 10(ΔTj = 123.4 °C).
Determine the average junction maximum temperature in repetitive avalanche
operation (Tj(AV)) using Equation 11 (Tj(AV) = 153 °C). Therefore, condition 2 issatisfied.
Finally based on the energy we can determine the limit number of operations atthese conditions from Figure 13, (No. events = 8e9)

Based on the above calculations, the operating conditions meet the repetitive
avalanche requirements with a limited number of repetitive events of 8e9.

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Power MOSFET single-shot and repetitive avalanche ruggedness rating

4

MOSFET & GaN FET Application Handbook

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113

Power MOSFET single-shot and repetitive avalanche ruggedness rating

4

114

Nexperia | A Power Design Engineer's Guide

8 References
1. Turn-Off Failure of Power MOSFETs — D.L. Blackburn, Proc. 1985 IEEE PowerElectronics Specialists Conference, pages 429 to 435, June 1985.
2. Power MOSFET failure revisited — D.L. Blackburn, Proc. 1988 IEEE PowerElectronics Specialists Conference, pages 681 to 688, April 1988.
3. Boundary of power-MOSFET, unclamped inductive-switching (UIS), avalanchecurrent capability — Rodney R. Stoltenburg, Proc. 1989 Applied Power 		Electronics Conference, pages 359 to 364, March 1989.

9 Figures
Figure 1. Unclamped inductive load test circuit for MOSFET ruggedness
evaluation........................................................................................................ 105
Figure 2. Gate-source voltage, VGS............................................................................... 107
Figure 3. Drain-source voltage, VDS.............................................................................. 107
Figure 4. Drain current, ID.............................................................................................. 107
Figure 5. Peak drain-source avalanche power, PDS(AL)M............................................. 107
Figure 6. Transient junction temperature profile of MOSFET during anavalanche event.............................................................................................. 107
Figure 7. BUK9K51-60R avalanche rating; avalanche current as a functionof avalanche time........................................................................................... 108
Figure 8. Gate pulse, VGS................................................................................................ 109
Figure 9. Drain-source voltage, VDS and repetitive drain- source avalanche ...............
current, IDS(AL)R................................................................................................. 109
Figure 10. Repetitive drain-source avalanche power, PDS(AL)R.................................... 110
Figure 11. Transient junction temperature components of MOSFET during ................
repetitive avalanche....................................................................................... 110
Figure 12. Repetitive avalanche rating; avalanche current........................................ 111
Figure 13. Repetitive avalanche rating; maximum number of avalanche
events as a function of avalanche energy.................................................. 111

4Power MOSFET single-shot and repetitive avalanche ruggedness rating

MOSFET & GaN FET Application Handbook

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Using RC Thermal Models

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Chapter 5
Using RC
Thermal Models
Application Note: AN11261

Using RC Thermal Models

5

MOSFET & GaN FET Application Handbook

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Nexperia | A Power Design Engineer's Guide

1 Introduction
Analysis of the thermal performance of power semiconductors is necessary toefficiently and safely design any system utilizing such devices. This chapter presents
a quick and inexpensive way to infer the thermal performance of power MOSFETs
using a thermal electrical analogy.
The thermal behaviour of power semiconductor devices can be predicted using RC
thermal models. The model types presented in this chapter are known as Foster
and Cauer models, consisting in networks of resistors and capacitors. Foster andCauer models are equivalent representations of the thermal performance of a
MOSFET and they can be used within a SPICE environment. This chapter provides
some basic theory behind the principle, and how to implement Foster and Cauer RC
thermal models. For convenience, Foster and Cauer RC thermal models arereferred to as RC models in the rest of this application note. Several methods ofusing RC thermal models, including worked examples, will be described.

2 Thermal impedance
RC models are derived from the thermal impedance (Zth) of a device (see Figure 1).
This figure represents the thermal behavior of a device under transient power
pulses. The Zth can be generated by measuring the power losses as a result ofapplying a step function of varying time periods.
A device subjected to a power pulse of duration > ~1 second, i.e. steady-state, hasreached thermal equilibrium and the Zth plateaus becomes the Rth. The Zthillustrates the fact that materials have thermal inertia. Thermal inertia means thattemperature does not change instantaneously. As a result, the device can handle
greater power for shorter duration pulses.
The Zth curves for repetitive pulses with different duty cycles, are also shown in
Figure 1. These curves represent the additional RMS temperature rise due to thedissipation of RMS power.
To assist this discussion, the thermal resistance junction to mounting base Rth(j-mb)
from the BUK7S1R0-40H data sheet, has been included in Table 1. The Zth in Figure
1 also belongs to the BUK7S1R0-40H data sheet.

Table 1: Steady state thermal impedance of BUK7S1R0-40H

118

SymbolParameter

Rth(j-mb)

thermal resistance fromjunction to mounting baseConditions Min
-

Typ

Max

Unit

0.35

0.4

K/W

1
Zth(j-mb)(K/W)δ = 0.5
10-1

0.2
0.1
0.05

10-2 0.02

single shot

P

δ=

tp
T

t

tp
T

10-3
10-6

10-5

10-4

10-3

10-2

10-1

tp (s)

1

Using RC Thermal Models

5

MOSFET & GaN FET Application Handbook

aaa-028930

Figure 1 | Transient thermal impedance from junction to mounting base as a function of pulse
duration for the BUK7S1R0-40H

3 Calculating junction temperature rise
To calculate the temperature rise within the junction of a power MOSFET, thepower and duration of the pulse delivered to the device must be known. If thepower pulse is a square, then the thermal impedance can be read from the Zthchart. The product of this value with the power gives the temperature rise within
the junction.
If constant power is applied to the device, the steady state thermal impedance canbe used i.e. Rth. Again the temperature rise is the product of the power and the Rth.
For a transient pulse e.g. sinusoidal or pulsed, the temperature rise within the
MOSFET junction becomes more difficult to calculate.
The mathematically correct way to calculate Tj is to apply the convolution integral.
The calculation expresses both the power pulse and the Zth curve as functions oftime, and use the convolution integral to produce a temperature profile (see Ref.
2).
τ
d
T
∫ P • = __ Z ( τ - t ) d t				[1]j(rise)

(t)

0

dt

th

However, this is difficult as the Zth(τ-t) is not defined mathematically.
An alternative way is to approximate the waveforms into a series of rectangularpulse and apply superposition (see Ref. 1).
While relatively simple, applying superposition has its disadvantages. The more
complex the waveform, the more superpositions that must be imposed to model

119

Using RC Thermal Models

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Nexperia | A Power Design Engineer's Guide

the waveform accurately.
To represent Zth as a function of time, we can draw upon the thermal electricalanalogy and represent it as a series of RC charging equations or as an RC ladder.
Zth can then be represented in a SPICE environment for ease of calculation of thejunction temperature.

4 Association between Thermal and Electricalparameters
The thermal electrical analogy is summarized in Table 2. If the thermal resistanceand capacitance of a semiconductor device is known, electrical resistances andcapacitances can represent them respectively. Using current as power, and voltage
as the temperature difference, any thermal network can be handled as an electrical
network.

Table 2: Fundamental parameters
Type

Resistance

Potential

EnergyCapacitanceElectrical(R = V/I)

R = resistance(Ohms)

V = PD (Volts)

I = current(Amps)

C = capacitance(Farads)

Thermal(Rth = K/W)

K = temperature
W = dissipatedRth = thermal
difference
power (Watts)
resistance (K/W)(Kelvin)

Cth = thermal
capacitance(thermal mass)

5 Foster and Cauer RC thermal models
Foster models are derived by semi-empirically fitting a curve to the Zth, the result of
which is a one-dimensional RC network Figure 2. The R and C values in a Foster
model do not correspond to geometrical locations on the physical device.
Therefore, these values cannot be calculated from device material constants as canbe in other modeling techniques. Finally, a Foster RC model cannot be divided orinterconnected through, i.e. have the RC network of a heat sink connected.
R1

R2

Rn

C1

C2

Cn
aaa-010334

Figure 2 | Foster RC thermal models

120Foster RC models have the benefit of ease of expression of the thermal impedance
Zth as described at the end of Section 2. For example, by measuring the heating or
cooling curve and generating a Zth curve, Equation 2 can be applied to generate afitted curve Figure 3:nZth(t) = ∑ i=1 Ri • [ 1 - exp ( - τit ) ]Where: τ i = R i • C i

			

[2]

				

[3]

The model parameters Ri and Ci are the thermal resistances and capacitances thatbuild up the thermal model depicted in Figure 2. The parameters in the analyticalexpression can be optimised until the time response matches the transient system
response by applying a least square fit algorithm.

Using RC Thermal Models

5

MOSFET & GaN FET Application Handbook

The individual expression, “i”, also draws parallels with the electrical capacitor
charging equation. Figure 3 shows how the individual Ri and Ci combinations, sum
to make the Zth curve.

Zth Curve
overlaid withsummed RC
Zth

RC ModelZth
RC curves,
representing
RC elementsTime

aaa-010335

Figure 3 | Foster RC thermal models

Foster models have no physical meaning since the node-to-node heat capacitances
have no physical reality. However, a Foster model can be converted into its Cauer
counter-part by means of a mathematical transformation (see Ref. 4).
An n-stage Cauer model can be derived from an n-stage Foster model and they will
be equivalent representations of the device thermal performance.
As seen for the Foster model, the Cauer Model also consists of an RC network but
the thermal capacitances are all connected to the thermal ground, i.e. ambient

121

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temperature as represented in Figure 4. The nodes in the Cauer Model can havephysical meaning and allow access to the temperature of the internal layers of thesemiconductor structure.
R2

R1
C1

C2

Rn
Cn

aaa-031516

Figure 4 | Cauer RC thermal models

Nexperia provides Foster and Cauer RC models for most of their Automotive PowerMOSFET products on the Product Information Pages. The models can be foundunder the Support tab, demonstrated below with BUK7S1R0-40H, see Figure 5.

Figure 5 | Nexperia RC thermal model documentation

Foster and Cauer RC thermal models allow application engineers to perform fastcalculations of the transient response of a package to complex power profiles.
In the following sections several examples of using RC thermal models will bepresented. Foster models and Cauer models are equivalent representations of thedevice thermal behaviour but in the described examples Cauer models will be used
as more representative of the physical structure of the device.
As shown in Figure 8, a schematic file is available on Nexperia website for
BUK7S1R0-40H Cauer model. Other products may not have this schematic fileavailable and may be provided with a netlist file for the Cauer network as shown in
Figure 6:122

*Part: BUK7S1R0-40H.subckt cauer 1 6 7
R1

1

2

0.00272144

R2

2

3

0.0220255

R3

3

4

0.00713124

R4

4

5

0.185679R5

5

60.182443

C1

1

79.29451e-05

C2

2

7

0.000514739

C3

3

7

0.00195047

C4

4

7

0.00305028

C5

5

7

0.0279554

Using RC Thermal Models

5

MOSFET & GaN FET Application Handbook.end cauer
Figure 6 | BUK7S1R0-40H Cauer model netlist

The netlist describes the same Cauer network as in Figure 8, and can be used tobuild the same schematic. Pin 1 in the netlist can be identified as the junction
temperature pin Tj in the schematic. Similarly pins 6, 7 as the Tamb pins in theschematic.
In order to simulate only the MOSFET pins 6 and 7 will both be tied to the ambient
voltage source, as shown in Figure. 8.
However, one of the advantages of using Cauer models is to allow to add external
networks to the MOSFET model, for example to model PCBs, heatsinks etc. In orderto do so pin 7 will be tied to ambient and pin 6 to the first pin of the external Cauer
network. For correct results, it is fundamental to make sure that the end pin of theexternal Cauer network is tied to the ambient source.

6 Thermal simulation examples
6.1 Example 1
RC thermal models are generated from the Zth curve. This example shows how towork back from an RC model and plot a Zth curve within a SPICE simulator. It allows
for greater ease when trying to read values of the Zth curve from the data sheet.
This and subsequent examples use the RC thermal model of BUK7S1R0-40H. Tmb
represents the mounting base temperature. It is treated as an isothermal and for123

Using RC Thermal Models

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Nexperia | A Power Design Engineer's Guide

this example it is set as 0 °C.
A single shot pulse of 1 W power is dissipated in the MOSFET. Referring to Figure 7;
for a single shot pulse, the time period between pulses is infinite and therefore theduty cycle δ = 0. Then the junction temperature Tj represents the transient thermal
impedance Zth.
P

tp

t
T

aaa-010337

Figure 7 | Single-shot pulse

Tj = Tmb + ΔT = 0 ºC + ΔT = ΔT
ΔT =P • Zth = 1 W • Zth

				

[4]

				

[5]

Equation 5 demonstrates that with P = 1 W, the magnitude of Zth equates to ΔT.
The following steps are used to set up and run simulations:
1. set up the RC thermal model of BUK7S1R0-40H in SPICE as shown in Figure 8
2. set the value of voltage source Vmb to 0, which is the value of Tmb
3. set the value of the current source I1 to 1
4. create a simulation profile and set the run time to 1 s
5. run the simulation
6. Plot the voltage at nod

124

5
BUK7S1R0-40H

Tj

BUK7S1R0-40H
Tamb

R1
0.00272144

Tamb

C2
0.000514739

R2
0.0220255

Tamb

C20.000514739

R2
0.0220255

TambR3
0.00713124

Tamb

R3
0.00713124

Tamb

R4
0.185679

Tamb

R40.185679

Tamb

R50.182443

C1
0.00009295

1
I1
I1

T

j
R1
0.00272144

C1
0.00009295

1

C30.0019547
C30.0019547

C4
0.00305028
C4
0.00305028
C5
0.0279554

TambC50.0279554

TambUsing RC Thermal Models

MOSFET & GaN FET Application Handbook

R5
0.182443

vmb
0
vmb
0

Tamb

.tran 1 Tuic
amb
.tran 1 uic

aaa-031522
aaa-031522

Figure 8 | BUK7S1R0-40H thermal model setup in SPICE

The simulation result in Figure 9 shows the junction temperature (voltage at Tj)
which is also the thermal impedance of BUK7S1R0-40H. The values of Zth atdifferent times can be read using the cursors on this plot within SPICE.

1
Tj
(°C)
10-1

10-2

10-3
10-6

10-5

10-4

10-3

10-2

10-1

t (s)

1

aaa-031523

Figure 9 | A plot of Tj from simulation

The value of the current source in this example is set to 1 A to represent 1 W
dissipating through the device. It can be easily changed to represent any value of
power. The simulation command can be changed for any duration to represent a
range of square power pulses.

125

Using RC Thermal Models

5

Nexperia | A Power Design Engineer's Guide

6.2 Example 2
Another method of generating the power profile, is to use measurements from theactual circuit. This information is presented to the SPICE simulation in the form of acomma-separated value (CSV) file giving pairs of time/power values. It can begenerated either as a summary of observations showing the points of change orfrom an oscilloscope waveform capture.
Two further methods of generating a power profile are discussed. One method isusing a PWL file. The other is to generate the power from an MOSFET electricalcircuit modeled in SPICE. The former is outlined first.
A source within a SPICE simulator can use a PWL file as an input. The contents of atypical PWL file is shown in Table 3. It can list the current, voltage or in thisexample, power over time. These files can be generated by typing values into a
spreadsheet editor and saving as a .csv file, or alternatively exporting waveforms
from an oscilloscope. The actual file itself should not contain any column headings.
To implement this procedure within a SPICE environment, follow the same steps as
described in Section 6.1 “Example 1”, but with the exceptions:
1. Set the property value of the current source to read from a PWL FILE and point it
to a .csv file for example: C:\Pulse file\filepulse.csv, which contains the power
profile listed in Table 3
2. Set the mounting base Tmb (Vmb) to 125 °C
3. Set the simulation run time to 0.6 s

126

5

MOSFET & GaN FET Application Handbook

Time (seconds)

Power (Watts)

0.000000

00.000001

120

0.004000120

0.004001

24

0.004002

24

0.100000

240.100001

240.10000280

0.200000800.20000280

0.200003

00.300000

00.30000180

0.315000800.315001240.400000

240.400001

00.500000

00.500001

120

0.515000120

BUK7S1R0-40H

Using RC Thermal Models

Table 3: Data example for use in a PWL file

Tj.tran 0.6 uic
PWL file=powerProfile.csv
I1

C1
0.00009295

R1
0.00272144

Tamb
C2
0.000514739

R2
0.0220255

TambC30.0019547

R3
0.00713124

TambC40.00305028

R4
0.185679

Tamb
C50.0279554

R5
0.182443

Tamb
vmb

Tamb

125

aaa-031526

Figure 10 | SPICE circuit implementing a PWL file with the thermal model of the BUK7S1R0-40H

127

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Nexperia | A Power Design Engineer's Guide

The simulation result is shown in Figure 11. The junction temperature and thermal
impedance values labeled in Figure. 11 demonstrate that the Zth value at 4 ms, andRth value, are in line with Figure 12. It represents the thermal impedance waveformshown in the BUK7S1R0-40H data sheet.
180
Tj
(°C)
150

180
P
(W)

Rth = 0.4 K/Wtemperature

150

120

120
power

90

90

60

60

30

30

0

0
0

0.1

0.2

0.3

0.4

0.5

180
Tj
(°C)
150

0.6

t (s)

a) simulation results

aaa-031527
180

Zth = 0.26 K/WP(W)

temperature

150

120

120

90

90

60

60
power

30

30

0

0

2

4

6

8

0
10

t (ms)b) first power pulse

aaa-031528

Figure 11 | Simulation results and first power pulse1
Zth(j-mb)(K/W)

Rth = 0.4 K/W

δ = 0.5

10-1

Zth = 0.26 K/W

0.2
0.1
0.05

10-2

single shot

0.02

P

δ=

T

t

tp
10-3
10-6

tp

T
10-5

10-4

10-3

10-2

10-1

tp (s)

1

aaa-028930

Figure 12 | Transient thermal impedance for BUK7S1R0-40H

The red lines highlight the thermal resistance and impedance for the example
shown in Figure 111286.3 Example 3
The aim of this example is to show how to perform thermal simulation using thepower profile generated from a MOSFET circuit.
Following the steps in Section 6.1, set up the thermal model of BUK7S1R0-40H, andset the mounting base temperature to 85 °C.
To set the power value in the current source, construct a MOSFET electrical circuitas provided in Figure 13. The power supply is 12 V. The gate drive supply is assigned
a value of 10 V. It is set to run for 50 cycles with a 1 ms period and a 50 % duty
cycle.

Using RC Thermal Models

5

MOSFET & GaN FET Application Handbook

The power dissipated in the MOSFET can be calculated from Equation 6 or for
greater accuracy; the gate current can be included into the calculation to give
Equation 7:
P = VDS • ID					[6]To improve accuracy:
P = VDS • ID + VGS • IG

					

[7]

The current source into the thermal model can now be defined as:
I = V(D) • I (VD) + V(G) • I (VG)					[8]Figure 13 demonstrates the link between the electrical circuit and the thermal
model circuit.
BUK7S1R0-40H.tran 50m uic.inc BUK7S1R0-40H.lib
Tj

1

V1

I = V(d)*I(Vd)+V(g)*I(Vg)

B1
I1

C1
0.00009295

R6
10 m

R1
0.00272144

12

Tamb

Vd
C2
0.000514739

R2
0.0220255

0

Tamb
C3
0.0019547

R3
0.00713124

TambC40.00305028

R4
0.185679

Tamb

Vg

BUK7S1R0-40H
d
Rg
g

0

10

M1

s

V2

PULSE (0 10 0 1u 1u 0.1m 1m 50)

C50.0279554

R5
0.182443

Tamb
Tamb

vmb1
0
85

aaa-031529

Figure 13 | SPICE circuit illustrating how to integrate an electrical circuit with a thermal model

129

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The resultant plot of Tj is shown in Figure 14. The maximum temperature of thejunction can once again be calculated from data sheet values by following the steps
outlined in Ref. 1.
180
Tj
(°C)
150

120

90

60

30

0

0

0.1

0.2

0.3

0.4

t (ms)

0.5

aaa-031530

Figure 14 | Inferred junction temperature (Tj) rise provided by Figure 12

7 Discussions
RC thermal models are not perfect. The physical materials used to build
Semiconductors have temperature-dependent characteristics. These characteristics
mean that thermal resistance is also a temperature-dependent parameter.
Whereas in Ohm’s law, the Ohmic resistance is usually considered to be constantand independent of the voltage. So the correspondence between electrical andthermal parameters is not perfectly symmetrical but gives a good basis for
fundamental thermal simulations.
In power electronic systems, the thermal resistance of silicon amounts to 2 % to
5 % of the total resistance. The error resulting from the temperature dependence
is relatively small and can be ignored for most cases. To obtain a more accurate
analysis, replace the passive resistors in the RC model with voltage-dependent
resistors. In these resistors, the change in temperature can correspond to change
in voltage.
A further limitation of the models presented is that the mounting basetemperature of the MOSFET Tmb, is set as an isothermal. This is rarely the case inreal applications where a rise in the mounting base temperature must be
considered. This rise is determined by calculating the temperature rise due to theaverage power dissipation (i.e. the heat flow) from the mounting base through toambient. It means that the models are of limited use for pulses greater than 1 s,
where heat begins to flow into the environment of the MOSFET. In this situation,
the thermal model for the MOSFETs, PCB, heat sink and other materials in
proximity must be included.

130

8 Summary
RC thermal models are available for Nexperia power MOSFETs on the Nexperia
website. The models can be used in SPICE or other simulation tools to simulate thejunction temperature rise in transient conditions. They provide a quick, simple andaccurate method for application engineers to perform the thermal design.

5
Using RC Thermal Models

MOSFET & GaN FET Application Handbook

RC thermal simulation of
power MOSFETs

9 References
1. Application note AN11156 - “Using Power MOSFET Zth Curves”. Nexperia
2. Application note AN10273 - “Power MOSFET single-shot and repetitive 		
avalanche ruggedness rating”. Nexperia
3. Combination of Thermal Subsystems Modeled by Rapid Circuit Transformation.
Y.C. Gerstenmaier, W. Kiffe, and G. Wachutka
4. JEDEC Standard JESD51-14 Transient Dual Interface Test Method for the
Measurement of the Thermal Resistance Junction-to-Case of Semiconductor
Devices with Heat Flow Through a Single Path131

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Nexperia | A Power Design Engineer's Guide

10 Tables
Table 1.
Table 2.
Table 3.

Steady state thermal impedance of BUK7S1R0-40H..............................118
Fundamental parameters.............................................................................120
Data example for use in a PWL file.............................................................127

11 Figures
Figure 1. Transient thermal impedance from junction to mounting baseas a function of pulse duration for the BUK7S1R0-40H..........................119
Figure 2. Foster RC thermal models.............................................................................120
Figure 3. Foster RC thermal models.............................................................................121
Figure 4. Cauer RC thermal models..............................................................................122
Figure 5. Nexperia RC thermal model documentation.............................................122
Figure 6. BUK7S1R0-40H Cauer model netlist...........................................................123
Figure 7. Single-shot pulse.............................................................................................124
Figure 8. BUK7S1R0-40H thermal model setup in SPICE.........................................125
Figure 9. A plot of Tj from simulation..........................................................................125
Figure 10. SPICE circuit implementing a PWL file with the thermal modelof the BUK7S1R0-40H....................................................................................127
Figure 11. Simulation results and first power pulse....................................................128
Figure 12. Transient thermal impedance for BUK7S1R0-40H...................................128
Figure 13. SPICE circuit illustrating how to integrate an electrical circuit .....................
with a thermal model.....................................................................................129
Figure 14. Inferred junction temperature (Tj) rise provided by Figure 12...............130

5
Using RC Thermal Models

MOSFET & GaN FET Application Handbook133

Designing RC snubbers

6

134

Nexperia | A Power Design Engineer's Guide

Chapter 6
Designing
RC snubbers

Designing RC snubbers

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MOSFET & GaN FET Application Handbook

Application Note: AN11160

135

Designing RC snubbers

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Nexperia | A Power Design Engineer's Guide

1 Introduction
This chapter describes the design of a simple “RC snubber circuit”. The snubber isused to suppress high-frequency oscillations associated with reverse recovery
effects in power semiconductor applications.

2 Test circuit
The basic circuit is a half-bridge and shown in Figure 1.
VDD

Q1

HS driverinductor

VCC
LS driverQ20V

aaa-002741

Figure 1 | The half-bridge circuit

Q1 and Q2 are BUK761R6-40E devices. The inductor could also be connected to 0 V
rather than VDD.
Inductor current is established in the red loop; Q2 is off and current is flowing
through Q1 body diode. When Q2 is turned on, current “commutates” to the blue
loop and the reverse recovery effect occurs in Q1. We observe the effect of Q1
reverse recovery on the VDS waveform of Q2; see Figure 2.

136

6
Designing RC snubbers

MOSFET & GaN FET Application Handbook

Q2
DSV(5V/div)
oscillation
frequency
31.25 MHzaaa-002742

Figure 2 | Reverse recovery-induced oscillation in Q2 VDS

The equivalent circuit is shown in Figure 3.

stray
inductance (LLK)

Q1
Coss
(CLK)

Q2

VDD

Q2
VDS
aaa-002743

Figure 3 | Equivalent circuit

We are primarily interested in the parasitic elements in the circuit:• LLK is the total stray or “leakage” inductance comprised of PCB trace inductance,
device package inductance, etc.
• The parasitic capacitance CLK is mainly due to Coss of the upper (Q1) device.
Q2 is treated as a simple switch. The oscillation can be eliminated (snubbed) byplacing an RC circuit across Q1 drain-source; see Figure 4137

6
Designing RC snubbers

Nexperia | A Power Design Engineer's Guide

stray
inductance (LLK)

Q1
Coss
(CLK)

Q2

RS
CS

VDD

Q2
VDS

aaa-002744

Figure 4 | Equivalent circuit with snubber components RS and CS

3 Determining CLK and LLK
Before we can design the snubber, we must first determine CLK and LLK. We could
attempt to measure CLK and LLK directly, but a more elegant method can be used.
For this LC circuit, we know that:							[1]where fRING0 is the frequency of oscillation without a snubber in place; see Figure 2.
If we add an extra additional capacitor across Q1 (Cadd), the initial oscillation
frequency from fRING0 to fRING1 (fRING1 < fRING0) will change. It can be shown that (seeSection 7 “Appendix A; determining CLK from Cadd, fRING0 and fRING1”):							[2]where:
							[3]So if we measure fRING0 (without Cadd), then add a known Cadd and measure fRING1,
we can determine CLK and LLK (two equations, two unknowns).
Cadd = 3200 pF was added in circuit, and fRING1 found to be 22.2 MHz (fRING0
previously found to be 31.25 MHz; see Figure 2).
from Equation 3:								[4]and from Equation 2:								[5]138

Rearranging Equation 1:
LLK =

1

[6]

(2π fRING0) 2 CLK

So with fRING0 = 31.25 MHz and CLK = 3239 pF:
1
LLK = _____________________________ = 8.01 x 10 -9 H = 8.0 nH(2 x π x 3.125 x 107) 2 x 3.239 x 10 -9[7]

and with fRING1 = 22.2 MHz and (CLK + Cadd) = 3239 pF + 3200 pF = 6439 pF:
1
LLK = ____________________________ = 7.98 x 10 -9 H = 8.0 nH(2 x π x 2.22 x 107) 2 x 6.439 x 10 -9

Designing RC snubbers

6

MOSFET & GaN FET Application Handbook[8]

In other words, the calculated value of LLK remains almost unchanged when we add
the additional 3200 pF capacitance. This is a good sanity check of the method for
determining CLK and LLK.

4 Designing the snubber - theory
If we replace CS in Figure 4 with a short-circuit, then we simply have the classic RLC
circuit found in text books. The response of this circuit to a step change in voltage(that is Q2 turning on) depends on the degree of damping (ζ or zeta) in the circuit;see Figure 5.
2.0
(1)
(2)
(3)
(4)

1.5

1.0

0.5

0

(5)
(6)
(7)

0

1

2

3

(1) ζ = 0.

(5) ζ = 0.7.

(2) ζ = 0.1.

(6) ζ = 1.

(3) ζ = 0.2.

(7) ζ = 2.

(4) ζ = 0.4.

4

aaa-002745

Figure 5 | Step response of an RLC circuit for various values of zeta (ζ)

139

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Nexperia | A Power Design Engineer's Guide

Designing RC snubbers

In theory the circuit oscillates indefinitely if ζ = zero, although this is a practicalimpossibility as there is always some resistance in a real circuit. As ζ increases
towards one, the oscillation becomes more damped that is, tends to decrease overtime with an exponential decay envelope. This is an “underdamped” response. The
case ζ = one is known as “critically damped” and is the point at which oscillation justceases. For values of greater than one (overdamped), the response of the circuit
becomes more sluggish with the waveform taking longer to reach its final value.
There is therefore more than one possible degree of damping which we could build
into a snubber, and choice of damping is therefore part of the snubber design
process.
For this configuration of RLC circuit, the relationship between ζ, RS, LLK and CLK is:
				

[9]

The snubber capacitor CS does not appear in Equation 9.
In some circuits, it is possible to damp the oscillations with RS alone. However, in
typical half-bridge circuits we cannot have a resistor mounted directly across Q1
drain source. If we did, then Q1 is permanently shorted by the resistor and thecircuit as a whole would not function as required. The solution is therefore to put
CS in series with RS, with the value of CS chosen so as not to interfere with normal
operation.
The snubber is a straightforward RC circuit whose cut-off frequency fC is:		1
F C = _______ 						 [10]2πRCCS
Again, we must choose which value of fC to be used, and there is no single correct
answer to this question. The cut-off frequency of the snubber must be low enoughto effectively short-circuit the undamped oscillation frequency fRING0, but not so low
as to present a significant conduction path at the operating frequency of the circuit(for example 100 kHz or whatever). A good starting point has been found to be
fC = fRING0.

140

5 Designing the snubber - in practice
We now have sufficient information to design a snubber for the waveform shownin Figure 2. To recap:
CLK = 3239 pF
LLK = 8.0 nH
fRING0 = 31.25 MHz								

[11]

1F C = _______ = fRING0		
2πRCCS			

[12]

			

Designing RC snubbers

6

MOSFET & GaN FET Application Handbook

The first task is to choose a value of damping (Figure 5). We have chosen ζ = 1, thatis, critical damping. Rearranging Equation 11 we have:
				

[13]

use 2 × 1.5 Ω in parallel to give 0.75 Ω.
Rearranging Equation 12 we have:							

[14]

use 4.7 nF + 2.2 nF to give 6.9 nF.
The snubber was fitted across Q1 drain source. The resulting waveform is shown in
Figure 6 together with the original (non-snubbed) waveform from Figure 2

a. Without snubber
Vertical scale is 2 V/div.

b. With snubber
aaa-002746

aaa-002747

Figure 6 | Q2 VDS waveform with and without snubber

141

Designing RC snubbers

6

Nexperia | A Power Design Engineer's Guide

As seen in Figure 6, the snubber has almost eliminated the ringing in the VDS
waveform. This technique could also be applied to the MOSFET in the Q2 position.

6 Summary• Reverse recovery effects in power devices can induce high frequency oscillations
in devices connected to them.
• A common technique for suppressing the oscillations is the use of an RC 		snubber.
• Design of an effective snubber requires the extraction of the circuit parasitic
capacitance and inductance. A method has been demonstrated for doing this.
• The snubbed circuit has been shown to be a variation on the classic RLC circuit.
• A method of determining values of snubber components has been 		demonstrated. The method has been shown to work well, using the example of
BUK761R6-40E MOSFETs

7 Appendix A; determining CLK from Cadd, fRING0
and fRING1
We know that:							

[15]

where fRING0 is the frequency of oscillation without a snubber in place and LLK and
CLK are the parasitic inductances and capacitances respectively.
If we add capacitor Cadd across Q1 drain-source, fRING0 is reduced by an amount “x”
where:
							

[16]

therefore							

[17]							

[18]							

[19]							

[20]							

142							

[21]							

[22]							

[23]

where:
							

[24]

Designing RC snubbers

6

MOSFET & GaN FET Application Handbook143

Designing RC snubbers

6

144

Nexperia | A Power Design Engineer's Guide

8 Figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.

The half-bridge circuit ................................................................................... 136
Reverse recovery-induced oscillation in Q2 VDS........................................ 137
Equivalent circuit ........................................................................................... 137
Equivalent circuit with snubber components RS and CS .......................... 138
Step response of an RLC circuit for various values of zeta (ζ) ................ 139
Q2 VDS waveform with and without snubber............................................ 141

Chapter 7
Half-bridge MOSFET
switching and its
impact on EMC
Application Note: AN90011

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

145Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

1 Introduction
Measuring and improving MOSFET switching behaviour to meet EMC requirements
and optimize reliability and efficency in half-bridge switching circuits.
Modern switching converters for low voltages (< 100 V) predominantly use power
MOSFETs as the switching devices.
Switching converter applications includes inverters to synthesise AC waveforms, orfor use in DC-to-DC converters. The switching devices are often arranged in a
simple half-bridge configuration. At some point in time one of the MOSFETs will be
actively switching (sometimes called the control FET in DC-to-DC applications) andthe other one will be switched off and acting as a diode during the commutation
event – this will be switched on once the switching event has finished and acts as asynchronous rectifier, (in DC-to-DC converters this MOSFET is referred to as a
syncFET).
The behaviour of the MOSFETs during the switching event strongly influences
efficiency and electromagnetic interference (emissions) goals.
In switch mode (PWM) designs, the switching behaviour of the MOSFET (or moregenerally the switching power devices) can influence the efficiency and theemissions from the system. This will apply to MOSFETs from any vendor, not justthose from Nexperia.
This application note describes a technique to measure the switching efficacy andvarious measures that can be taken to improve the switching behaviour, thus
meeting efficiency, EMC requirements and reliability goals.

2 Double pulse testingThis is a relatively simple method of determining the switching behaviour of a pairof MOSFETs in a half-bridge switching circuit. MOSFET Q1 is switched off and acts
only as a diode, MOSFET Q2 is actively switched on and off twice: hence "double
pulse testing". The behaviour at a particular current level and DC voltage level canbe analysed. This testing is useful for comparing MOSFETs from different suppliers,
since the detailed design – the technology - of the devices is unlikely to be exactly
the same. Figure 1 shows the simplified schematic diagram of the double pulse test
circuit.

146

+ve
power

Q1
Inductor

Gate
drive

Q2

RsenseCurrent
monitor-ve
power
aaa-031171

Figure 1 | Simplified schematic diagram of the double pulse test circuit

Simplified switching behaviour is shown in Figure 2
60 V

V(sw_mode)
Switch node voltage

Turn off

Turn on

+ve
power

40 V
20 V
0V
0 µs
15 V

5 µs

10 µs

15 µs

20 µs

25 µs

30 µs

35 µs

40 µs

V(driver)
Driver signal

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

Gate
drive

10 V
5V
0V
0 µs
75 A

5 µs

10 µs

15 µs

20 µs

25 µs

30 µs

35 µs

40 µs

25 µs

30 µs

35 µs

40 µs

I(Rsense)

Current
monitor-ve
power

Low-side drain current

50 A
25 A
0A
0 µs

5 µs

10 µs

15 µs

20 µs

aaa-031198

Figure 2 | Simplified switching behaviour

Three waveforms are shown in Figure 2: the top one shows the switch node
voltage, the middle one shows the gate driver waveform to the low-side MOSFET
and the lower waveform shows the current in the low-side device. The regions ofinterest are at turn-off of the low-side switch and then at turn-on for the second
pulse. Assuming the gap between the first pulse and second pulse is relatively short(but long enough to ensure proper switch off of the switched device before itis switched on again) then the current will be maintained at the target level since
the diode will present a voltage of around 1 V to the inductor, so di/dt will be low.
The MOSFET gate driver can be to the user’s choice however Nexperia uses a highcurrent driver so that the external gate drive resistor strongly determines the147Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guideswitching speed, rather than this being influenced by the gate driver IC itself.

2.1 Measurement system
Current sensing is a critical consideration. A high bandwidth sensor is required. A
coaxial shunt is used in Nexperia’s test circuit since the bandwidth can be morethan 1 GHz however the additional circuit inductance is in the region of 5 nH. Other
methods of measuring current are possible such as Rogowski coils, Hall Effectprobes or current transformers but these also suffer from various compromises
either in bandwidth or significant additional circuit inductance. Some specialist
Rogowski based probes can reach ~80 MHz bandwidth. The goal is to havesufficient bandwidth in the current sensor whilst minimising the additional circuit
inductance. It may also be a requirement to have isolation of the probe signal fromthe circuit (especially if double pulse testing is performed on high voltage systems).
Here are a few tips regarding the measurement system: the current rise time may
be in the area of up to 5 ns, so the current signal bandwidth would be 70 MHz,BW ≈ 0.35/trise, see ref (1). A voltage rise time of 5 ns would also be of 70 MHz.
The probe BW should be 3x to 5x signal BW to take into account the 3 dB at thequoted BW. The oscilloscope should therefore have an analogue bandwidth thesame or higher than the probes when used with x10 voltage probes (1 MΩ input
impedance, note that the nominal oscilloscope bandwidth might quoted for a
50 Ω input impedance). The probes should be correctly compensated before use.
The sampling rate needs to be at least 2.5x the analogue bandwidth for reasonable
resolution. In this example the probes and oscilloscope would need to be 200 MHz
analogue BW minimum and the sampling rate would need to be 500 Msamples/s
minimum.
Note also that it can take some time for signals to travel along probe cables so
the voltage probes and current probe cables should be the same length or thedifference in time delay compensated for: a cable difference of 0.5 m could
introduce a time difference of around 4 ns. This could have a significant effect on
the measurement of switching energy. It is always worth spending a little time tomake sure that the measurement system is correctly set up for accuracy, see ref (1).
In double pulse testing, the waveforms of interest are low-side VDS (switch node
voltage measured as close to the MOSFET terminals as possible), high-side VDS (maybe measured using a differential probe method using high-side VD minus low-side
VDS ), low-side VGS and low-side ID. Ideally high-side VGS should also be measured.

148

2.2 MOSFET switching simulation
SPICE simulation is used to produce some example waveforms. The BUK7S1R0-40H
is used for most of this chapter, except where otherwise stated. The simulation
circuit is an approximate model of the physical double pulse test circuit withparasitic elements added, see Figure 3. Nexperia has found that there can be agood correlation between SPICE simulation and physical measurements of thereal circuit, as long as the device models are accurate (as is the case for Nexperia
MOSFET models) and the circuit is realistically modelled, see Appendix A.
Figure 4 shows the MOSFET turn-off event and Figure 5 shows the MOSFET turn-on
event.
L20

Rg1

L25

1n
Lload

HsG

1n

(RG external)

L21
1n

L22
1n
LsD
L26

(RG external)

1n

L11

LsG

1n
M2
BUK7S1R0-40H

Double pulse

L14

1n
L1
(ceresl)

M1
BUK7S1R0-40H

HsS

Rg2

L12

L10

HsD

1n

L2
(ceresl)
m=3

1n

L4
(ceresl)m=3

L5(elcoesl)

C5

C1

C2

C3

C4

(cerval)(cerval)m=3(cerval)(cerval)m=3

R1(ceresr)

R2(ceresr)

R3(ceresr)

R4(ceresr)

Vp

L16

1n
L3
(ceresl)

L6(elcoesl)m=3

L7(elcoesl)

C6

C7(supply)m=3

R6(elcoesr)
m=3

L13

L15

L17

1n

1n

1n

R7
(elcoesr)

Rsupply

V3

C8

m=3

R5
(elcoesr)

L8(elcoesl)
m=3

R8(elcoesr)
m=3

Vn

L23
1n

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

RsenseL24
5n

I

aaa-031139

Figure 3 | SPICE simulation circuit for double pulse test circuit

(V)

60

90

I(Rsense)

50
40

30
0

30

V(HsD, HsS)

20

-30

V(LsG)

10-60

V(LsD)

0

-90
-120

-10
-20

(A)

60

13.2

13.4

13.6

13.8

14.0

14.2

14.4

14.6

t (µs)

14.8-150

V(LsG) = low-side MOSFET gate to source voltage
I(Rsense) = low-side MOSFET drain current
V(HsD, HsS) = high-side MOSFET drain to source voltage V(LsD) = low-side MOSFET drain to source voltage
aaa-031185

Figure 4 | Turn-off waveform149

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

(V)

60

90

I(Rsense)

50
40

30
0

30

V(HsD, HsS)

20

-30

V(LsG)

10-60

V(LsD)

0

-90
-120

-10
-20

(A)

60

13.2

13.4

13.6

14.0

13.8

14.4

14.2

14.6

t (µs)

14.8-150

V(LsG) = low-side MOSFET gate to source voltage
I(Rsense) = low-side MOSFET drain current
V(HsD, HsS) = high-side MOSFET drain to source voltage V(LsD) = low-side MOSFET drain to source voltage
aaa-031185

Figure 5 | Turn-on waveform

2.3 MOSFET turn-off waveform description
These waveforms require some explanation as to why they look the way they do.
Reference is made to MOSFET capacitances, see Figure 7. Consider first the
turn-off waveform in Figure 6. This is divided into 6 time periods t0 to t5.

(V)

60

90

I(Rsense)

50
40

30
0

30

V(HsD, HsS)

20

-30

V(LsG)

10-60

V(LsD)

0

-90
-120

-10
-20

(A)

60

t0
13.2

13.4

t1

t2

13.6

13.8

t3

t4
14.0

t5
14.2

14.4

14.6

t (µs)

14.8-150

V(LsG) = low-side MOSFET gate to source voltage
I(Rsense) = low-side MOSFET drain current
V(HsD, HsS) = high-side MOSFET drain to source voltage V(LsD) = low-side MOSFET drain to source voltage
aaa-031188

Figure 6 | MOSFET turn-off waveforms, showing reference time periods

During period t0, the low-side device is on and the high-side device is off. This is a
steady state condition. During time period t1, the device turn-off process begins as
the gate driver removes charge from the gate capacitance (CISS = CGS + CGD, see
Figure 7) of the device and the gate- source voltage starts to fall. VDS rises very150

slightly as the RDS(on) of the device begins to increase, in accordance with the“drain-source on-state resistance as a function of gate-source voltage”
characteristic found in Nexperia MOSFET data sheets.

CGD
NMOS
CDS
CGS

aaa-022333

Figure 7 | MOSFET internal capacitances

During t2, once VGS falls to the minimum value required to sustain the output
inductor current (as determined by the output characteristic, see Figure 8), VGS isapproximately constant (this is the "Miller Plateau" and the VGS value = Vp). During
this time the gate driver current discharges CGD (causing VGS to fall and VDG to rise),until the switch node voltage has risen to a voltage that allows the diode to
conduct current. At this point VGS can fall to its threshold voltage (during t3) as
current is commuted from the low-side MOSFET to the high-side MOSFET body
diode.
400
ID
10 V
(A)
320

7V

6V

Ohmic region

240

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

VGS= 5.5 V

saturation region
5V

160

80

4.5 V
4V

0

0

1

2

3
VDS (V)4

aaa-030487

Figure 8 | Example of MOSFET output characteristics

151Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

Towards the end of t2 and moving into t3, the low-side MOSFET VDS reaches thelevel of the DC supply voltage and continues to increase. At this point the body
diode of the high-side device can start to turn on and the ID of the low-side MOSFET
can commutate to the high-side device.
The rate at which this happens is now controlled by the gate driver discharging Cissas in period t1 and the incremental slope of the MOSFET transfer curve(transconductance) of the MOSFET (the transfer curve can be found in Nexperia
Power MOSFET data sheets). The resulting rate of change of drain current dID/dt
causes a voltage to be produced across all circuit inductances affected by thechange in ID. Referring to Figure 3, the inductances affected by dID/dt include L10,
L11, L20, L21, L22, L23 and L24, as well as any component package inductances (forexample L1, L2 and others not shown in Figure 3). This is known as the loop
inductance. This results in the large voltage overshoot seen on the low-side
MOSFET VDS, the overshoot is 14.5 V. The peak value corresponds to the peakdID/dt. The equal change of current in the high-side MOSFET source IS results in asmall negative voltage spike across the high-side device due to L20 and L21:
note that the measuring point for the voltages includes some circuit impedancecontribution in this example and that the MOSFET model includes packageparasitics.
This also adds to the voltage spike observed on the low-side MOSFET. The total
circuit inductance from the capacitors through both MOSFETs is often referred toas the loop inductance. The sum of the layout (PCB) inductances is 10 nH in Figure 3
however another 1 nH comes from the device internal inductances, then there are
additional inductances coming from the DC link capacitors and layout which are notso simple to describe. The peak dID/dt is 1.1 A/ns in Figure 4 so 12.1 V is due to the
inductances and the voltage overshoot is 13.0 V. Note that the high-side body
diode is conducting, so this accounts for about 0.7 V of the difference.
At the end of t3 and start of t4, ID in the low-side MOSFET reaches zero. All thecurrent has transferred to the high-side MOSFET body diode. At this point, thevoltage across the low-side MOSFET is still more than 30 V given a supply voltage of
20 V. Oscillation is inevitable and a simple circuit model as shown in Figure 9
illustrates the situation.

152

Open switch
after short time(NC)

Rdamping(NC)

Initialvoltage= 12 V

COSS

Lparasitic

aaa-031172

Figure 9 | Oscillation at MOSFET turn-off

COSS is the low-side MOSFET effective capacitance (CDS in parallel with CGD, CGS isshorted out by the gate driver). This is VDS dependant and the value at 32 V is usedin this example). Lparasitic is the loop inductance. If COSS is averaged to 1.6 nF and
Lparasitic is 11 nH then the oscillation frequency will be 37.9 MHz. This is quite close
to the observed oscillation frequency around 35 MHz. The damping resistancewhich is due to the device construction and resistance due to the layout will reduce
the frequency and non-linearity of COSS results in a slightly bigger effective
capacitance being present.

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

In period t5, the MOSFET turn-off switching transition is complete. The low
frequency oscillation is due to resonance of the ceramic capacitors with the ESL ofthe electrolytic capacitors and layout inductance. The resonant frequency isapproximately 2 MHz in this example. The ceramic capacitors are 100 nH and the
ESL of the electrolytic capacitor is 20 nH in the simulated example, on its own theresonant frequency is calculated at of 3.6 MHz. Factoring in the layout inductancewould account for the discrepancy. See Section 3.2.1 for more details about low
frequency ringing.
The switching loss in the low-side MOSFET at turn-off, (over the t2 and t3 time
periods), is given by:∫V(LsD) * I(Rsense)dt

			

[1]

Where V(LsD) = low-side MOSFET drain to source voltage. The switching loss in thehigh-side MOSFET is negligible.

153

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

2.4 MOSFET turn-on waveform description
A typical turn-on waveform is shown in Figure 10. In time period t0 the low-side
MOSFET is off. Current is flowing in the body diode of the high-side MOSFET.
During t1 the gate voltage on the low-side MOSFET starts to rise. Nothing happens
until the threshold voltage is reached.
(V)

60

90

50

60

40

30

I(Rsense)30

0

V(LsD)

20

-30
-60

10

V(LsG)

0-90

V(HsD, HsS)

-10
-20

(A)

t1

t0
20.8

20.9

21.0

t2
21.1

-120

t5
t3

t4

21.3

21.4

21.5

t (µs)

21.6-150

I(Rsense) = low-side MOSFET drain current
V(LsG) = low-side MOSFET gate to source voltage
V(LsD) = low-side MOSFET drain to source voltage V(HsD, HsS) = high-side MOSFET drain to source voltage
aaa-031189

Figure 10 | MOSFET turn-on waveforms, showing reference time periods

At the start of time period t2, the gate voltage continues to rise until significant
current starts to flow. The rate of change of drain current dID/dt increases and thenbecomes more constant, as defined by the transfer characteristic (see Nexperia
MOSFET data sheet) dID/dt is also dependant on gate driver current in Ciss. The
circuit inductances start to interact with the MOSFET. Notice how the MOSFET VDS
falls and then plateaus at around 5 V in this example. This corresponds with thehighest dID/dt observed, interacting with the loop inductance (see Section 2.3).
dID/dt is therefore limited by the supply voltage and the loop inductance, rather
than the MOSFET capability.
Another limitation to dID/dt is due to the device source and layout inductance tothe connection of the gate drive (represented by L23 in Figure 3). Packages with asource clip such as LFPAK will have a higher dID/dt capability compared to wire
bonded packages such as DPAK due to the package source inductance. Notice also
the high-side VDS shows a “hump” due to dID/dt interacting with L20 and L21 (seeFigure 3). VGS as observed at the pins of the low-side device is also affected due toinductance of the source leg, the actual VGS applied to the MOSFET die will be fairly
constant due to the MOSFET gate capacitance CGS, however the voltage across the
package source inductance is added. This can just be observed in Figure 10. At theend of period t2 the low-side drain current reaches the load inductance current. The
low-side VDS doesn’t fall at this point. This is because the high-side body diodemust be turned off first. Note that the low-side MOSFET is experiencing a relatively
high current and also a relatively high VDS, hence there will be some switching loss154

associated with this condition.
At the start of time period t3, the current in the low-side MOSFET increases beyond
the load current. The high-side MOSFET body diode is being switched off, anegative current is flowing in it and the depletion layer inside the body diode isforming in order to support reverse voltage, charge is being stored in the diodejunction. This can be considered as a capacitance (sometimes referred to as the
diffusion capacitance Cd, the associated charge may be referred to as Qs). The
low-side drain current approaches its peak value.
Once the depletion layer begins to form, the high-side VDS will begin to increase.
This happens at the start of period t4. The low-side VDS will fall. The low-side draincurrent will still increase a little further before reaching a peak, partly due tocharging the diffusion capacitance Cd but also due to charging the junction
capacitance Cj in the high-side body diode. The pn junction can be considered as aparallel plate capacitor, with the distance between the “plates” containing thedepletion region as the dielectric material. There comes a point where the current
flowing into the depletion region becomes relatively small and the current flowing
into the junction capacitance dominates. This is just before the peak current isobserved in the low-side MOSFET. The junction capacitance characteristic(especially with respect to VDS) now determines the rate of change of current
observed in the body diode and low-side MOSFET. It also determines the rate ofrise of the high-side VDS. This is unlike the situation seen during turn-off, where thegate driver and CGD control the dVDS/dt, now this is controlled by the energy stored
in the loop inductance and COSS . The charge stored in COSS (CJ = CDS, COSS ≈ CDS +CGD) is referred to as QOSS. The process of switching off the high-side MOSFET body
diode during time periods t3 and t4 is known as reverse recovery and is described in
ref (5) and other references.

7
Half-bridge MOSFET switching and its impact on EMC

MOSFET & GaN FET Application Handbook

The high-side VDS will usually exceed the nominal DC supply level due to the dID/dt
and loop inductance. This is referred to as a VDS “spike” and can be problematic.
Notice also that the observed low-side VGS waveform shows some oscillations. VDS
oscillations are coupled into the MOSFET gate directly as a result of the capacitive
divider formed by CGD and CGS (see Figure 7). Secondly, the observed waveform isaffected by source lead and layout inductance, represented by L23.
In period t5, a high-frequency decaying oscillation is observed superimposed on alower frequency DC link oscillation. The high-frequency oscillation is due to theresonance of the high-side COSS and the loop inductance Lparasitic. The low frequency
oscillation is due to the resonance of the DC link ceramic capacitors and inductanceof the electrolytic capacitors and circuit layout as discussed in the MOSFET turn-off
condition, (see Section 3.2.1). The oscillations decay over time.
The switching loss in the low-side MOSFET at turn on, (over the t2, t3 and t4 time
periods.), is given by:155

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide∫V(LsD) * I(Rsense)dt[1]

The switching loss in the high-side MOSFET, (over the t4 time period), is given by:∫V(HsD, HsS) * (I(Rsense)-I(Lload))dt[2]

3 Methods to improve switching performance
In Section 2, the switch on and switch off transients were explored using double
pulse testing. In the waveforms presented, some undesirable effects can be seensuch as the large voltage spike on the high-side VDS when the low-side MOSFET
switches on, also some oscillations at turn-on and turn-off. This is a cause of
unwanted noise which may propagate to the outside world and may interfere withthe operation of radio systems but may also cause disturbances in other analogue
and digital systems (hence the use of the terms such as electromagnetic
compatibility or electromagnetic interference: EMC / EMI).

3.1 Voltage spikes
VDS spikes are observed in both the switch off and switch on waveforms. At
low-side turn on, it is possible that the high-side VDS might experience a spike which
may reach or exceed the rating of the MOSFET. It is good practice to limit this spiketo < 80% of the MOSFET rating to achieve good reliability and also to keep the
amplitude of the following oscillations quite small.
The following sections consider some methods to minimise these voltage spikes.

3.1.1 Circuit layout
The loop inductance was described in Section 2.3 and Section 2.4. This total
inductance should be minimised as far as possible by using compact layout
methods, so that the loop area is reduced: if the loop is considered as a turn of acoil then a large coil diameter will result in a larger self inductance. Furthermore,
the magnetic field generated by the loop can interfere with nearby wiring
according to Faraday’s law.

156

+ve supply

ground plane(on another PCB layer)

2.07 nH
ground plane(on another PCB layer)+ve supply

4 mm

switch node
2.52 nH
1.55 nH

3.5 nH

4.04 nH

switch node

4.75 nH

2.07 nH

VDC
I/P

D1

3.5 nHC

1.55 nH

VDC

S2

2.52 nHaaa-031121

PCB design 1: loop inductance = 10.3 nH

4.04 nH

C

I/P

4.75 nH

D1

S2

aaa-031122

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

PCB design 2: loop inductance = 8.2 nH

Figure 11 | Half-bridge layout possibilities showing inductances

A layout arrangement as shown in Figure 11 helps to reduce the loop inductance.
Design 1 has a loop inductance of 10.3 nH compared to design 2 with a loop
inductance of 8.1 nH, according to simulation (ignoring device inductances). Usingsmaller devices can help to reduce the loop area further.
It was seen in Section 2 that switching behaviour depends on the gate drivervoltage and impedance, the package layout source inductance in the gate driverloop and the loop inductance.

3.1.2 Reducing dI/Dt
In the case of the low-side MOSFET switch off waveform (Figure 4, Figure 6), thevoltage spike depends on the loop inductance and the dI/Dt. Since the current isfalling, the parasitic inductance will add to the supply voltage as seen at the drain
terminal of the MOSFET. The ringing was due to the step voltage change when thedrain current (and dI/Dt) reaches zero. This can be achieved by slowing the gate
drive by some method such as slew rate control at the driver IC, increasing gate

157

Half-bridge MOSFET switching and its impact on EMC

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resistance or adding capacitance between gate and source. This will increase theswitching time and therefore the switch off loss.

ID
(A)

trr
dI/dt

ta

tb

0
0.25 IRMtb
S= tat rr = t a + t b

dIR/dtIRM

t (s)Figure 12 | Reverse recovery waveform definitions

In the case of the low-side MOSFET switch on, it was seen in Figure 10 that there isa significant VDS spike across the high-side MOSFET which is acting as a diode. The
spike is due to the loop inductance and dIR/dt (see Figure 12) which cannot bedirectly controlled but which is a function of the dI/dt that is controllable. The time
ta is approximately constant and it depends on the time it takes for the depletion
region to form in the pn junction of the body diode (i.e. recombination time). A
higher dI/dt will therefore result in a higher IRM value and higher energy in the loop
inductance.
dI/dt and dIR/dt are related by the softness factor of the body diode (tb/ta), which isreasonably constant over a range of dI/Dt values. dIR/dt multiplied by the loop
inductance causes a voltage overshoot. This can be absorbed by the MOSFET
output capacitance COSS. This capacitance is non-linear with respect to VDS, (seeFigure 13), which doesn’t help with keeping VDS spikes under control. COSS defines
the shape of the tb region (and consequently the dIR/dt) and is a characteristic of
the MOSFET technology.

158

20,000
COSS(pF)
16,00012,000

8,000

4,000

0

0

5

10

15

20

25

30
35
VDS (V)

aaa-031120

Figure 13 | COSS as a function of VDS

Adding additional capacitance external to the MOSFET can help to “linearise” thecapacitance and reduce the VDS spike. This is the capacitance part of an RC snubber.

3.2 Ringing

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

This occurs because a stimulus is applied to a resonant circuit. In the half-bridge,
inductances are usually attributed to the circuit layout and parasitics within thecomponents, usually the loop inductance. Capacitances are attributed to the
MOSFETs (e.g. COSS) and bypass capacitors

3.2.1 Bypass capacitors
Bypass capacitors should be placed as close to the half-bridge as possible (tominimise the loop area as discussed in 3.2.1), ideally using surface mount devices.
The value should be chosen such that resonance will be well damped. A 100 nF 100
V C0G capacitor has an ESL of approximately 1.1 nH and an ESR of 20 mΩ. A 1 μF
100 V X7R capacitor has similar ESL and ESR. This means that the 1 μF capacitor has
a better damping capability compared to the 100 nF capacitor, since thecharacteristic impedance will be closer in value to the capacitor's ESR. The C0G is amore stable dielectric however a higher capacitance is more beneficial. It was
shown that the 2 MHz oscillation is due to the capacitance of the ceramic bypass
capacitors and the inductance of the layout and electrolytic capacitors. This is also
borne out in practice.
In the example simulation circuit, 100 nF capacitors are compared with 1 μF
capacitors, see Figure 14.

159

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

45

45

VDS
40
(V)

VDS
40
(V)

35

35

30

30

25

25

20

20

15

15

10

10

5

5

0

0

-5
20.5

21.5

22.5

Cbypass = 100 nF

t (µs)

23.5

aaa-031191

-5
20.5

21.5

Cbypass = 1 μF

22.5

t (µs)

23.5

aaa-031192

Figure 14 | Effect of bypass capacitor selection: 100 nF vs. 1 μF

The 2 MHz oscillation is shifted to a lower frequency and is a lower amplitude, theESR and PCB track resistance is providing damping. However the higher frequency
oscillation of around 35 MHz remains and is changed only slightly.

3.2.2 Adding a snubber
By adding an RC snubber across the MOSFET drain and source, oscillations can be
minimised. AN11160 (chapter 6 in this book) - Designing RC snubbers (ref 3) covers
this topic. Another good reference is (ref 4) which proposes a method to optimise
the snubber capacitor. The snubber will attenuate the unwanted high frequency
oscillations as shown in Figure 15. A 10 nF + 2.2 Ω snubber is applied to thehigh-side and low-side devices. VDS for the high-side device is shown at low-side
turn-on (i.e. high-side body diode turn-off). This also damps the oscillations at
low-side turn-off. SPICE programs allow a sweep of parameters to give an indicationof the snubber values which are likely to give the desired outcome.

160

45

35
VDS
(V) 30

VDS
40
(V)
35

25

30

20

25
20

15

15

10

10

5

5

0

0
-5
20.8

21

21.2

21.4

t (µs)

-5
20.8

21.6

without snubber

21

21.2

21.4

with snubbert (µs)

21.6

aaa-031193

Figure 15 | Effect of adding a snubber

3.3 Feedback into the gate signal
There is a risk of feedback into the gate signal due to high dVDS/dt. If this exceeds
the gate threshold voltage (VGS(th)) then a significant shoot through current canflow in the half-bridge. This is due to the MOSFET capacitances CGD, CDS and theexternal gate drive resistance forming a potential divider. If a shoot throughcurrent occurs, this will significantly increase the switching losses. This is illustrated
in Figure 16, see ref (7).
i=C

dV
dt

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

18

MOSFET

VGS

RG

CGS

CDS
VGS

Voltage (V)

16
CGD

L

MOSFET

14

10
8

HS VGS (V)6

LS VGS (V)4

Switch Node (V)20Some of this charge is sunk by the driver but some will
charge CGS and VGS will increase. Both MOSFETs canpotentially be on at the same time causing a shortcircuit and current to shoot through the MOSFETs

VGS “bounces
back upduring the dV/dt

dV/dt across the highside FET causes acapacitive current tocharge CGD

12

0

2

4

6

Time (µs)

8aaa-026552

Figure 16 | Dynamic behaviour of a MOSFET half-bridge

Additional external capacitance is used to control dV/dt. Crss is relatively small andnon-linear giving rise to gate glitches and sudden shoot through current. The
external capacitance would be much larger in value and quite linear, producing acontrolled dV/dt. See section 3.4.2.

161

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7Nexperia | A Power Design Engineer's Guide

3.4 Switching speed control
3.4.1 Effect of varying RG
Varying the external gate resistance will affect switching losses in the low-side
MOSFET; see Figure 17, Figure 18 and Figure 19. In this case, measurements havebeen made using the BUK7S1R5-40H.
160
E
(mJ)
120

turn-off energy

80
turn-on energy
40

0

0

20

40

60

80

100

RG

120

aaa-031195

Figure 17 | Effect of varying RG on switching loss (example values only)

6
di/dt(A/ns)

dIr/dt4

2

turn-on di/dt

turn-off di/dt
0

0

20

40

60

80

100

RG

120

aaa-031196

Figure 18 | Effect of varying RG on di/dt, dIr/dt and turn-off di/dt (example values only)

162

12
dV/dt
(V/ns)
turn-on dV/dt
8

4

turn-off dV/dt
0

0

20

40

60

80

100

RG

120

aaa-031197

Figure 19 | Effect of varying RG on turn-on dV/dt and turn-off dV/dt (example values only)

Increasing RG will increase switching loss but the benefit is better controlledswitching and improvements in spiking and ringing. There may also be an increased
risk of gate bounce so it may be necessary to add a small capacitor between gate
and source.
For best performance, RG for turn-on should be large than RG for turn-off. di/dt is
controlled by the VGS value around the Miller plateau voltage Vp:

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

IG = (VGS - Vp) / RG for example:
IG(on) = (10 - 4.3) / 10 = 0.57 A
IG(off) = (0 - 4.3) / 10 = -0.43 A
Hence turn on dI/Dt may be different to turn off dI/Dt depending on the value of RG.
dI/Dt can also be controlled by carefully selecting the location of the gate drive
ground connection point, this is equivalent to adjusting L23 in Figure 3.

3.4.2 Effect of adding external gate to source capacitanceIn case of gate bounce risk, (see Section 3.3), an external CGS can be added toreduce the high frequency impedance of the gate driver whilst still having relatively
high RG The effect on switching speed (dI/Dt) is relatively modest, as seen below in
Figure 20. This is for a BUK7S1R0-40H with an RG external value of 22 Ω.

163

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

1000
dI/Dt
turn-on

900

800
turn-off
700

600

500

0

2

4

6

8

CGS (nF)10

aaa-031498

Figure 20 | CGS effect on MOSFET dI/Dt

3.4.3 Effect of adding gate to drain capacitance
Adding some drain voltage feedback via a gate – drain capacitor would reduce
dV/dt without affecting dI/Dt and will reduce the voltage spike at turn-off. Losses
are increased. Reducing dV/dt is beneficial because in a motor drive, the switch
node voltage is presented directly to the motor windings. The motor windings canact as antennae so that the frequency domain content of the waveform can radiate
into the environment. The highest frequency component of the waveform will beapproximately 0.35/rise time or 0.35/fall time.
160
dV/dt
120

80

40

0

0

1

2

3

4

CGD (nF)

5

aaa-031499

Figure 21 | External CGD effect on controlling MOSFET turn-off

Figure 21 shows the effect of adding a capacitor to a BUK7S1R0-40H with a series
resistor of 10 Ω. Usually a series resistor RGD is added to CGD in order to prevent
oscillations, see Figure 22. The value of CGD can be in the range of 1 – 5 nF for a

164

device such as BUK7S1R0-40H, much bigger than the value of the MOSFET internal
Crss. The value of RGD can be in the range of 10 – 50 Ω for this device, higher values
start to diminish the effect of having the external CGD. Note that Figure 22 applies
to the low-side controlling switch in the case of the double pulse test circuit.
In the case of the MOSFET acting as a synchronous rectifier then the effect ofadding external CGD will reduce dv/dt however the value of dV/dt can be muchhigher. See Figure 23.

RGD
CGD

aaa-031501

Figure 22 | Adding CGD and RGD

6000

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

dV/dt

4000

2000

0

0

1

2

3

4

QGD (nF)

5

aaa-031502

Figure 23 | External CGD effect on the synchronous rectifier MOSFET

It is recommended that some SPICE simulation is performed first before
experimenting to determine the optimum values of external resistors andcapacitors.

165

Half-bridge MOSFET switching and its impact on EMC

7Nexperia | A Power Design Engineer's Guide

4 Impact on EMC and efficiency
The significant voltage transients and oscillations which have been described in
Section 2 are the source of electromagnetic interference. This must be suppressedin order to meet CISPR25 based requirements for automotive and CISPR 11 and thelike for industrial, telecoms and consumer equipment. These kinds of standards
consider conducted and radiated emissions. How do these transient voltages andcurrents cause interference? In the next paragraphs, emissions from digital circuits
will be ignored however the same basic principles apply, the difference being thatthe voltages are much lower than in the power circuit and the frequency rise andfall times are much faster.
In an ideal PWM controlled half-bridge MOSFET circuit, it is still necessary to takecare of EMI by using screening and filters. This is because simple analysis of the halfbridge waveforms feature rectangular blocks of current with fast rising and fallingedges from the DC link and the output switch node will feature fast rising andfalling voltages. See Figure 24.
itvt
Vehiclebattery

3 or more
phasesDifferential
mode filter

aaa-031173

Figure 24 | Sources of electromagnetic noise in a half-bridge circuit

5 Conclusions
This chapter is intended to give the reader a reasonably detailed understanding of
how the half-bridge switching circuit is a source of electromagnetic noise andpower loss. In particular, the reverse recovery characteristic of the MOSFET body
diode can be a key factor to achieving good EMC behaviour and lower losses. Sometechniques to reduce switching noise at source have been considered and a brief
overview of how the noise propagates to the outside world have been considered.

166

Benefits of low Qrr
MOSFETs in motor
control applications

Benefits of low Qrr
MOSFETs in switching
applications

Visit blog
Qrr overlooked andunderappreciated inefficiency battle

Developing reliable
MOSFETs how difficult
can it be

Visit blog
Automotive Trench 9Power MOSFETs designed
for performance andendurance

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

6 References
1. High Speed Digital Design – a Handbook of Black Magic, Howard Johnson &Martin Graham
2. Power and small-signal MOSFET frequently asked questions and answers
TN00008, section 7.8, Nexperia
3. Designing RC snubbers” AN11160, Nexperia
4. Optimising snubber design through frequency-domain analysis”, EDN
5. Semiconductor device modelling with SPICE 2nd Edition, Giuseppe Massobrio &Paulo Antognetti
6. EMC for product designers, Tim Williams
7. Designing in MOSFETs for safe and reliable gate-drive operation AN90001,
Nexperia

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7Nexperia | A Power Design Engineer's Guide

7 Appendix A: Comparison between SPICE
simulation and practical measurements for the
BUK7S1R0-40H in a double pulse test circuit
The SPICE circuit used is as shown in Figure 3. The following values were used in thesimulation: ceresr = 20 m, ceresl = 1.13 n, cerval = 100 n, cerval1 = 100 n,
elcoesr = 40 m, elcoesl = 20 n.
Both turn-on and turn-off conditions are considered.
SPICE model is modified from standard: TT parameter is approximately half of
published value (to give better reverse recovery alignment with double pulse
measurements).
The following test conditions were applied: RG = 30 Ω, IL = 60 A, VDC = 20 V.

(V)

60

90

50

60
30

40

I(Rsense)30

0

V(LsD)

20

-30
-60

10

V(LsG)

0-90

V(HsD, HsS)

-10
-20

(A)

20.8

20.9

-120
21.0

21.1

I(Rsense) = low-side MOSFET drain current
V(LsD) = low-side MOSFET drain to source voltage

21.2

21.3

21.4

21.5

t (µs)

21.6-150

V(LsG) = low-side MOSFET gate to source voltage
V(HsD, HsS) = high-side MOSFET drain to source voltage
aaa-031186

Figure 25 | SPICE turn-on waveform

168

(V)

60

90

50

60

40

(A)

30

I(Rsense)30

0

V(LsD)

20

-30
-60

10

V(LsG)

0-90

V(HsD, HsS)

-10

-120
-150

-20
14.0

14.1

14.2

14.3

14.4

14.5

14.6

14.7

t (µs)

14.8

I(Rsense) = low-side MOSFET drain current
V(LsG) = low-side MOSFET gate to source voltage
V(HsD, HsS) = high-side MOSFET drain to source voltage V(LsD) = low-side MOSFET drain to source voltage
aaa-031238

Figure 26 | Measurement turn-on waveform

(V)

60

90

I(Rsense)

50
40

30
0

30

V(HsD, HsS)

20

-30

V(LsG)

10-60

V(LsD)

0

-90
-120

-10
-20

(A)

60

Half-bridge MOSFET switching and its impact on EMC

7

MOSFET & GaN FET Application Handbook

13.2

13.4

13.6

13.8

I(Rsense) = low-side MOSFET drain current
V(LsD) = low-side MOSFET drain to source voltage

14.0

14.2

14.4

14.6

t (µs)

14.8-150

V(LsG) = low-side MOSFET gate to source voltage
V(HsD, HsS) = high-side MOSFET drain to source voltage

aaa-031185

Figure 27 | SPICE turn-off waveform

169

Half-bridge MOSFET switching and its impact on EMC

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Nexperia | A Power Design Engineer's Guide

(V)

60

90

I(Rsense)

50

(A)

60
30

40

0

30

V(HsD, HsS)

20

-30

V(LsG)

10-60

V(LsD)

0

-90
-120

-10

-150

-20
11.0

11.2

11.4

11.6

11.8

12.0

12.2

12.4

12.6

t (µs)

I(Rsense) = low-side MOSFET drain current
V(LsG) = low-side MOSFET gate to source voltage
V(HsD, HsS) = high-side MOSFET drain to source voltage V(LsD) = low-side MOSFET drain to source voltage
aaa-031239

Figure 28 | Measurement turn-off waveform

Table 1: Meausrement and SPICE simluation values
Turn-on(measured)Turn-on(SPICE)

Turn-off(measured)Turn-off(SPICE)

dI/Dt

1.12 A/ns

1.12 A/ns

dI/Dt

0.82 A/ns

1.05 A/ns

dIr/dt2.96 A/ns

ΔIRM

10.09 A

2.78 A/ns

dV/dt

0.31 V/ns

0.45 V/ns

12.2 A

V(LsD)

29 Vta

34.2 V

11.43 ns12.8 ns

-

-

-

tb

5.17 ns

7.2 ns

-

-

-

dV/dt

4.54 V/ns

5.7 V/ns---

V(HsD, HsS)

27 V

41.2 V---8 Tables
Table 1.

Measurement and SPICE simulation values............................................... 170

7
Half-bridge MOSFET switching and its impact on EMC

MOSFET & GaN FET Application Handbook

171

Half-bridge MOSFET switching and its impact on EMC

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9 Figures
Figure 1. Simplified schematic diagram of the double pulse test circuit.............. 147
Figure 2. Simplified switching behaviour.................................................................... 147
Figure 3. SPICE simulation circuit for double pulse test circuit .............................. 149
Figure 4. Turn-off waveform......................................................................................... 149
Figure 5. Turn-on waveform.......................................................................................... 150
Figure 6. MOSFET turn-off waveforms, showing reference time periods............ 150
Figure 7. MOSFET internal capacitances..................................................................... 151
Figure 8. Example of MOSFET output characteristic................................................ 151
Figure 9. Oscillation at MOSFET turn-off.................................................................... 153
Figure 10. MOSFET turn-on waveforms, showing reference time periods............. 154
Figure 11. Half-bridge layout possibilities showing inductances............................ 157
Figure 12. Reverse recovery waveform definitions..................................................... 158
Figure 13. COSS as a function of VDS................................................................................ 159
Figure 14. Effect of bypass capacitor selection: 100 nF vs. 1 μF............................... 160
Figure 15. Effect of adding a snubber........................................................................... 161
Figure 16. Dynamic behaviour of a MOSFET half-bridge............................................ 161
Figure 17. Effect of varying RG on switching loss (example values only)................. 162
Figure 18. Effect of varying RG on turn-on dI/Dt, dIr/dt and turn-off dI/Dt(example values only)..................................................................................... 162
Figure 19. Effect of varying RG on turn-on dV/dt and turn-off dV/dt(example values only)..................................................................................... 163
Figure 20. CGS effect on MOSFET dI/Dt.......................................................................... 164
Figure 21. External CGD effect on controlling MOSFET turn-off............................... 164
Figure 22. Adding CGD and RGD........................................................................................ 165
Figure 23. External CGD effect on the synchronous rectifier MOSFET..................... 165
Figure 24. Sources of electromagnetic noise in a half-bridge circuit....................... 166
Figure 25. SPICE turn-on waveform............................................................................... 168
Figure 26. Measurement turn-on waveform................................................................ 169
Figure 27. SPICE turn-off waveform............................................................................... 169
Figure 28. Measurement turn-off waveform................................................................ 170

Chapter 8
Failure signature ofelectrical overstresson power MOSFETs
Application Note: AN11243

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1 Introduction
When Power MOSFETs fail, there is often extensive damage. Examination of thesize and location of the burn mark, the failure signature, provides information
about the type of fault condition which caused the failure. This chapter provides acatalogue of failure signatures from common electrical overstress failure modes.
The catalogue can be used in forensic investigation of the underlying root cause of
failure to improve module design and reliability.
Power MOSFETs are used to switch high voltages and currents, while minimizing
their own internal power dissipation. Under fault conditions however, it is possible
to apply voltage, current and power exceeding the MOSFET capability. Fault
conditions can be either due to an electrical circuit failure or a mechanical fault
with a load such as a seized motor. This leads to Electrical Overstress (EOS).
Typically the consequence of EOS is the short circuiting of at least 2 of the 3
MOSFET terminals (gate, drain, source). In addition, high local power dissipation in
the MOSFET leads to MOSFET damage which manifests as burn marks, die crack
and in extreme cases as plastic encapsulation damage.
Examination of the size and location of the burn mark, the failure signature,provides information about the type of fault condition which caused the failure.
Common fault conditions are:• ElectroStatic Discharge (ESD)• Unclamped Inductive Switching (UIS) - commonly called Avalanche or 		
Ruggedness• Linear Mode operation• Over-current
Packaged MOSFETs have been deliberately destroyed under these conditions.
Images recorded of the ensuing burn marks on the silicon surface, provide a‘Rogue’s Gallery’ to aid the explanation of EOS failures.
Section 1.1 to Section 1.4 gives an overview of the common failure signatures.
Appendices in Section 2.1 to Section 2.17 provide further images.

1741.1 ESD - Human body model
1.1.1 EOS method
ESD pulses were applied using a standard Human-body Model ESD circuit; fordetails see AEC - Q101 - REV - May 15, 1996. Voltage of the applied pulse wasprogressively increased until device failure was observed.
R1

S1

1500 Ω

HIGH VOLTAGE
PULSE
GENERATOR

C1
100 pFS2

terminal A

DUT
SOCKET

short

R2
500 Ω

terminal B
aaa-004855

Figure 1 | Typical circuit for Human body Model ESD simulation

1.1.2 Fault condition simulated
Human body model ESD simulates situations when a voltage spike is applied to the
MOSFET exceeding the maximum voltage that can be sustained by the gate oxideof either gate-source or gate-drain. The pulse is applied with 1500 Ω series
resistance between the voltage origin and the MOSFET, which limits the rate of riseof the MOSFET gate voltage. Either human handling, electrical test equipment or
malfunctioning circuits can easily apply such voltage pulses.

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1.1.3 Signature
Failure site is found in an edge cell of the MOSFET structure. Outer edge cells andcells near the gate are the first to be subjected to the incoming voltage pulse andare thus the first sites where the voltage exceeds the gate-oxide capability. The
signature differs from Machine Model failures in that the fail site does not showsuch a strong tendency to group near the gate, due to the slower rise in gate
voltage.

175

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Table 1: Examples of Human Body Model ESD failure signature
Device name

BUK9508-55A

Cell pitch
(μm)

Image

Comments
Fail site is gate
oxide of edge cell;see Section 2.1“Human bodymodel EOS of
BUK9508-55A”
for further images

9 (hexagon)

aaa-004856

BUK9Y40-55B

Fail site is gate
oxide of edge cell;see Section 2.2“Human bodymodel EOS ofBUK9Y40-55B”
for further images

4 (stripe)

aaa-004857

PSMN011-30YL

Fail site is gate
oxide of edge cell;see Section 2.3“Human bodymodel EOS of
PSMN011-30YL”
for further images

2 (stripe)

aaa-004858

PSMN8R5-100PSF

BUK7Y3R0-40H

176

2.5 (stripe)

1.5 (stripe)

Fail site is gate
oxide of edge cell,see Section 2.4"Human bodymodel EOS ofPSMN8R5-100PSF"
for further images.

Fail site is gate
oxide of edge cell,see Section 2.5"Human bodymodel EOS ofBUK7Y3R0-40H"
for further images.

1.2 Unclamped Inductive Switching (UIS)(Avalanche or Ruggedness)
1.2.1 EOS method
Inductive energy pulses were applied using a standard UIS circuit; for details see
AEC - Q101-004 - REV - May 15, 1996. A fixed inductance value is selected. Current
in the inductor prior to switching the MOSFET was progressively increased until
device failure was observed.

Table 2. UIS ruggedness test circuit and waveforms
Vbr
BVdss
Iav
drain
I
gate
voltage

VDD

L

L

gate
source0tav
VDD

aaa-004859

Figure 2 | Circuit diagram for UIS ruggedness test

Vg
tg(on)

aaa-004860

Figure 3 | Waveforms obtained from UIS test

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1.2.2 Fault condition simulated
UIS simulates situations when a MOSFET is switched off in a circuit in which there is
inductance. The inductance can be deliberate (such as an injector coil in a diesel
engine system), or parasitic. As the current cannot decay to zero instantaneously
through the inductance, the MOSFET source-drain voltage increases to take thedevice into avalanche breakdown. The energy stored in the inductance is thendissipated in the MOSFET.

1.2.3 Signature
Failure site is found in an active MOSFET cell. The burn-mark is usually round in
shape, indicating a central failure site and subsequent thermal damage.
If the avalanche event is long in duration (~ ms), then burn marks locate at central
sites on the die, where there is maximum current flow and reduced heat

177

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dissipation. The sites are often adjacent to wire bonds/clip bonds where current
density is high, but not directly under the wire bond/clip bond as it provides a local
heat sink. Failure is at the hottest location of the die.
For short avalanche events (~ μs), the burn marks can take on more random
locations over the die surface. The temperature rise in the chip is more uniformwith negligible chance for current crowding and local heating on these time scales.
For even shorter avalanche events, the burn marks can locate at die corners due tothe discontinuity in cell structure at these locations.

Table 3: Examples of Unclamped Inductive Switching failure signature
Device name

BUK7L06-34ARC

Cell pitch
(μm)

Image

Comments

Round burnin active area;
see Section 2.6“Unclamped
inductive
switching EOS ofBUK7L06-34ARC”
for further images

9 (hexagon)

aaa-004861

BUK9Y40-55B

Round burnin active area;
see Section 2.7“Unclamped
Inductive
Switching EOS ofBUK9Y40-55B”
for further images

4 (stripe)

aaa-004862

PSMN7R0-30YL

2 (stripe)

aaa-004863

178

Round burnin active area;
see Section 2.8“Unclamped
inductive
switching EOS ofPSMN7R0-30YL”
for further images

Device name

PSMN8R5-100PSF

BUK7Y3R0-40H

Cell pitch
(μm)

Image

Comments

2.5 (stripe)

Round burnin active area;
see Section 2.9"Unclamped
inductive
switching EOS ofPSMN8R5-100PSF"
for further images.

1.5 (stripe)

Round burnin active area;
see Section 2.10
"Unclamped
inductive
switching EOS ofBUK7Y3R0-40H"
for further images.

1.3 Linear mode operation
1.3.1 EOS method
A Safe Operating Area (SOA) graph is included in all power MOSFET data sheets.
Outside the defined safe region, the power dissipated in the FET cannot be
removed, resulting in heating beyond the device capability and then device failure.
MOSFETs were taken and a fixed source-drain voltage applied. Current pulses ofdefined duration were applied and the current was increased until MOSFET failure
was observed.

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103
ID
(A)

limit RDS(on) = VDS / ID

102

tp = 10 s

100 s

10

1
DC
10-1
1

10

VDS (V)

1 ms
10 ms
100 ms102

aaa-004864

Figure 4 | Safe operating area; continuous and peak drain currents as a function of drain-source
voltage179

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1.3.2 Fault condition simulated
Linear mode operation is common during device switching or clamped inductive
switching and is not a fault condition unless the SOA is exceeded. Linear mode EOS
simulates situations when a MOSFET is operated in Linear mode for too long. Thissituation can also occur if, when intending to turn the FET on, the gate signal
voltage to the FET is too low. This condition can also arise when intending to holdthe FET in the Off-state with high drain-source voltage. If the gate connection is
lost, the gate voltage capacitively rises and the same Linear mode fault condition
occurs

1.3.3 Signature
The hottest location of the die is a failure site that is usually at central sites on thedie. The center of the die is where there is maximum current flow and reduced heatdissipation. The sites are often adjacent to wire bonds/clip bonds where current
density is high, but not directly under the wire bond/clip bond as it provides a local
heat sink.

Table 4: Examples of linear mode failure signature
Device name

BUK7L06-34ARC

Cell pitch
(μm)

Image

Comments
Burns located incenter of die
adjacent towire-bonds;
see Section 2.11
“Linear mode
EOS ofBUK7L06-34ARC”
for further images

9 (hexagon)

aaa-004865

BUK9Y40-55B

Burn adjacent
to locationof clip bond incenter of die;see Section 2.12
“Linear mode
EOS ofBUK9Y40-55B”
for further images

4 (stripe)

aaa-004866

180

Device name

PSMN7R0-30YL

Cell pitch
(μm)

Image

Comments

Burn adjacent
to locationof clip bond incenter of die;see Section 2.13
“Linear mode
EOS of
PSMN7R0-30YL”
for further images

2 (stripe)

aaa-004867

PSMN8R5-100PSF

BUK7Y3R0-40H

2.5 (stripe)Burn between
source bond wires
in activearea,
see Section 2.18
"Linear mode
EOS of
PSMN8R5-100PSF"
for further images

1.5 (stripe)Burn close to thesource clip locationin the active
area,
see Section 2.19
"Linear mode
EOS ofBUK7Y3R0-40H"
for further images

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1.4 Over-current
1.4.1 EOS method
The maximum current-handling capability is specified on the data sheet for PowerMOSFETs. This capability is based on the current handling capability of wires orclips, before which fusing will onset, combined with the ability to dissipate heat.
Exceeding this rating can result in catastrophic failure.

181

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IDM

drain current

peak drain current

VGS = 10 V; Tmb = 100 °C; see Figure 1

-

53

A

VGS = 10 V; Tmb = 25 °C; see Figure 1

-

76

A

tp ≤ 10 µs; pulsed; Tmb = 25 °C; see Figure 3

-

260

A
aaa-005071

Figure 5 | Example of maximum current rating from the data sheet of PSMN7R0-30YL

1.4.2 Fault condition simulatedOver-current occurs if a FET is turned on with no element in the circuit to limit thecurrent, resulting in a supply voltage being applied fully over the drain-source
terminals of the FET. Typically this occurs if a load has been short-circuited.
Alternatively if 2 FETs are operating in a half-bridge, over-current can ensue if bothare turned on together.

1.4.3 Signature
Failure site is initially where the current handling connections (wires or clips) meetthe die. Normally damage is extensive however in over-current conditions, andspreads over the entire die surface with evidence of melted metallization andsolder joints.
For wire-bonded packages, there is often evidence of fused wires. For clip-bonded
packages, die crack is commonly observed.

Table 5: Examples of over-current failure signature
Device name

BUK7L06-34ARC

182

Cell pitch(μm)

9 (hexagon)

Image

Comments
Burns located incenter of die
adjacent towire- bonds.
Secondary damageof remelted top
metal and solder
die attach;see Section 2.14
“Over-current
aaa-004868
EOS ofBUK7L06-34ARC”
for further images

Device name

PSMN7R0-30YL

Cell pitch
(μm)

Image

Comments
Burn adjacent tolocation of clip
bond in centerof die;see Section 2.15
“Over-current
EOS of
PSMN7R0-30YL”
for further images

2 (stripe)

aaa-004869

2 Appendices2.1 Human Body Model EOS of BUK9508-55A
Table 6: Human body model EOS
BUK9508-55A

Cell structure:
Package:
Die size:
EOS condition:

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MOSFET & GaN FET Application Handbook

9 mm hexagons
TO-2205.5 mm x 4.5 mm
5 kV HBM pulse

aaa-004869

aaa-004899
Fails located in edge cells, distributed around edgeof device183

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aaa-004900

Figure 6 | Sample image 4; after AI removal

aaa-004901

Figure 7 | Sample image 4; after AI removal,
close-up

aaa-004902

Figure 8 | Sample image 19; after AI removal

aaa-004903

Figure 9 | Sample image 19; after TEOS removal,
close-up2.2 Human Body Model EOS of BUK9Y40-55B
Table 7: Human body model EOS
BUK9Y40-55B

Cell structure:
Package:
Die size:
EOS Condition:

4 μm stripe
LFPAK (clip bond)
2.5 mm x 1.35 mm
450 V to 650 V HBM
pulse

aaa-004904

Fails located randomly over die with increased
grouping in edge cells. Some fails subjected to ATE
testing to create additional damage to highlight
fail site

aaa-004905

Figure 10 | Sample image 5; after AI removals

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aaa-004906

Figure 11 | Sample image 5; after TEOS removal,
close-up

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2.3 Human Body Model EOS of PSMN011-30YL
Table 8: Human body model EOS
PSMN011-30YL

Cell structure:
Package:
Die size:
EOS Condition:

2 μm stripe
LFPAK (clip bond)
1.7 mm x 1.2 mm
200 V to 210 V HBM
pulse

aaa-004907

Fails located in edge cells

aaa-004908

Figure 12 | Device 4; after AI removals

		

aaa-004910

Figure 14 | Device 5; after AI removals

186

aaa-004909

Figure 13 | Device 7; after AI removal

aaa-004911

Figure 15 | Device 10; after AI removal

2.4 Human Body Model EOS ofPSMN8R5-100PSF
Table 9: Human body model EOS
PSMN8R5-100PSF

Cell structure:
Package:
Die size:
EOS Condition:

2.5 μm stripe
SOT78
4 mm x 2.67 mm
1.4 kV to 1.8 kV HBM
pulse

Figure 16 | Device 4 after AI removals

Figure 18 | Device 5 after TEOS removal

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Figure 17 | Device 7 after AI removal		

Figure 19 | Device 10 following decapsulation

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2.5 Human Body Model EOS of BUK7Y3R0-40H
Table 10: Human body model EOS
BUK7Y3R0-40H

Cell structure:
Package:
Die size:
EOS Condition:

1.5 μm stripe
SOT6692.65 mm x 2.15 mm
2 kV HBM pulse

Figure 20 | Device 2 after AI, barrier and TEOS etch

Figure 21 | Device 3 after AI, barrier
and TEOS etch

Figure 22 | Device 4 after AI, barrier and TEOS etch188

2.6 Unclamped Inductive Switched EOS ofBUK7L06-34ARC
Table 11: Unclamped inductive switching EOS
BUK7L06-34ARC
Cell structure:
Package:
Die size:
EOS Condition:

9 mm hexagons
Small round burn marks, randomly distributed
TO-220 (clip bond)over active area, close to but not directly
4.3 mm x 4.3 mm
0.2 mH; 80 A to 110 A under wire-bonds

aaa-004912

Figure 23 | Sample image 1

aaa-004913

Figure 24 | Sample image 2

aaa-004914

Figure 25 | Sample image 3

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MOSFET & GaN FET Application Handbook

aaa-004915

Figure 26 | Sample image 4189

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2.7 Unclamped Inductive Switched EOS ofBUK9Y40-55B
Table 12: Unclamped inductive switching EOS
BUK9Y40-55B

Cell structure:
Package:
Die size:
EOS Condition:

4 μm stripe
LFPAK (clip bond)
2.5 mm x 1.35 mm
Red dots: 0.1 mH,
76 A to 80 A
Yellow dots: 15 mH,
7 A to 9 A
aaa-004916

Small round burn marks, randomly distributed overactive area, close to but not directly under clip
bondaaa-004917

Figure 27 | Sample image 41; 0.1 mH		

aaa-004919

190

Figure 29 | Sample image 51; 15 mH		

aaa-004918

Figure 28 | Sample image 43; 0.1 mH

aaa-004920

Figure 30 | Sample image 55; 15 mH

2.8 Unclamped Inductive Switched EOS ofPSMN7R0-30YL
Table 13: Unclamped inductive switching EOS
PSMN7R0-30YL

Cell structure:
Package:
Die size:
EOS Condition:

2 μm stripe
LFPAK (clip bond)
2.5 mm x 1.35 mm
Red dots: 0.1 mH,
48 A to 51 A
Yellow dots: 15 mH,
16 A to 18 A
aaa-004921

Small round burn marks, randomly distributed overactive area, close to but not directly under clip
bondaaa-004922

Figure 31 | Sample image 6; 0.1 mH		

aaa-004924

Figure 33 | Sample image 18; 3.5 mH

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MOSFET & GaN FET Application Handbook

aaa-004923

Figure 32 | Sample image 8; 0.1 mH

aaa-004925

Figure 34 | Sample image 20; 3.5 mH

191

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2.9 Unclamped Inductive Switched EOS ofPSMN8R5-100PSF
Table 14: Unclamped inductive switching EOS
PSMN8R5-100PSF

Cell structure:
Package:
Die size:
EOS Condition:

2 μm stripe
SOT78
4 mm x 2.67 mm
Teal dot - 25 mH
Orange dot - 100 uH

Figure 35 | Device 1 upper (orange) hotspot

Figure 36 | Device 1 lower (orange) hotspot

Figure 37 | Device 4 upper (orange) hotspot

Figure 38 | Device 4 lower (orange) hotspot

Figure 39 | Device 6 (teal) hotspot

Figure 40 | Device7 (teal) hotspot

Figure 39 | Device 6 (teal) hotspot

Figure 40 | Device 7 (teal) hotspot

Figure 41 | Device 8 (teal) hotspot

Figure 42 | Device 9 (teal) hotspot

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2.10 Unclamped Inductive Switched EOS ofBUK7Y3R0-40H
Table 15: Unclamped inductive switching EOS
BUK7Y3R0-40H

Cell structure:
Package:
Die size:

1.5 μm stripe
SOT6992.65 mm x 2.15 mm

Figure 43 | Device1

Figure 45 | Device 3

194

		

Figure 44 | Device 2

2.11 Linear mode EOS of BUK7L06-34ARC
Table 16: Linear mode EOS
BUK7L06-34ARC
Cell structure:
Package:
Die size:
EOS condition:

9 mm hexagon
TO-220 (clip bond)
4.3 mm x 4.3 mm
15 V, 3 A Burn marks located in middle of the die adjacent to wire bonds
30 V, 1.5 A Burn mark and location are more discrete at 20 V, 1.5 A

aaa-004926

Figure 46 | Sample image 1: 15 V, 3 A		

aaa-004928

Figure 48 | Sample image 3: 15 V, 3 A		

aaa-004927

Figure 47 | Sample image 2: 15 V, 3 A

Failure signature of electrical overstress on power MOSFETs

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aaa-004929

Figure 49 | Sample image 4: 15 V, 3 A

195

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aaa-004930

Figure 50 | Sample image 1: 30 V, 1.5 A		

aaa-004932

Figure 52 | Sample image 3: 30 V, 1.5 A		

aaa-004931

Figure 51 | Sample image 2: 30 V, 1.5 A

aaa-004933

Figure 53 | Sample image 4: 30 V, 1.5 A

2.12 Linear mode EOS of BUK9Y40-55B
Table 17: Linear mode EOS
BUK9Y40-55B

Cell structure: 4 μm stripe
Package:
LFPAK (clip bond)
Die size:
2.5 mm x 1.35 mm
EOS condition: 20 V, 3.5 A, 30 ms
20 V, 3 A, 60 ms
30 V, 1.4 A, 60 ms
aaa-004916

Burn marks in center of die, adjacent but notdirectly under clip bondaaa-004935

Figure 54 | Sample image 61; 20 V, 3.5 A, 30 ms

aaa-004937

Figure 56 | Sample image 63; 20 V, 3.5 A, 30 ms

Failure signature of electrical overstress on power MOSFETs

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MOSFET & GaN FET Application Handbook

aaa-004936

Figure 55 | Sample image 62; 20 V, 3.5 A, 30 ms

aaa-004938

Figure 57 | Sample image 64; 20 V, 3.5 A, 30 ms

197

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aaa-004937

Figure 58 | Sample image 66; 20 V, 3 A, 60 ms		

aaa-004942

Figure 60 | Sample image 68; 20 V, 3 A, 60 ms

aaa-004944

Figure 62 | Sample image 71; 30 V, 1.4 A, 60 ms

198

aaa-004938

Figure 59 | Sample image 67; 20 V, 3 A, 60 ms

aaa-004943

		Figure 61 | Sample image 69; 20 V, 3 A, 60 ms

aaa-004946

Figure 63 | Sample image 72; 30 V, 1.4 A, 60 ms

aaa-004945

aaa-004947

Figure 64 | Sample image 73; 30 V, 1.4 A, 60 ms

Figure 65 | Sample image 74; 30 V, 1.4 A, 60 ms

2.13 Linear mode EOS of PSMN7R0-30YL
Table 18: Linear mode EOS
PSMN7R0-30YL
Cell structure:
Package:
Die size:
EOS condition:

2 μm stripe
LFPAK (clip bond)
2.3 mm x 1.35 mm
0.1 mH, 48 A to 51 A
3.5 mH, 16 A to 18 A

Burn marks in center of die, adjacent but notdirectly under clip bondaaa-004948

Figure 66 | Sample image 1; 15 V, 2.5 A, 100 ms

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MOSFET & GaN FET Application Handbook

aaa-004951

		

Figure 67 |Sample image 2; 15 V, 2.5 A, 100 ms

199

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aaa-004953

Figure 68 | Sample image 4; 15 V, 2.5 A, 100 ms

aaa-004955

Figure 70 | Sample image 11; 15 V, 5 A, 1 ms

aaa-004957

Figure 72 | Sample image 13; 15 V, 5 A, 1 ms

200

aaa-004954

Figure 69 | Sample image 5; 15 V, 2.5 A, 100 ms

aaa-004956

		

Figure 71 | Sample image 12; 15 V, 5 A, 1 ms

aaa-004958

Figure 73 | Sample image 14; 15 V, 5 A, 1 ms

2.14 Linear mode EOS of PSMN8R5-100PSF
Table 19: Linear mode EOS
PSMN8R5-100PSF

Cell structure: 2.5 μm stripe
Package:
SOT78
Die size:
4 mm x 2.67 mm
EOS condition: Teal dot - 50 V
10 ms pulse length
Orange dot - 70 V
1 ms pulse length

Figure 74 | Device 5 upper (orange) hotspot

Figure 75 | Device 6 lower (orange) hotspot

Figure 76 | Device 7 upper (orange) hotspot

Figure 77 | Device 8 lower (orange) hotspot

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Figure 78 | Device 1 (teal) hotspot

Figure 79 | Device 2 (teal) hotspot

Figure 80 | Device 3 (teal) hotspot

Figure 81 | Device 4 (teal) hotspot

2.15 Linear mode EOS of BUK7Y3R0-40H
Table 20: Linear mode EOS
BUK7Y3R0-40H

Cell structure: 1.5 μm stripe
Package:
LFPAK (clip bond)
2.3 mm x 1.35 mm
Die size:
EOS condition: 0.1 mH, 48 A to 51 A
3.5 mHm 16 A to 18 A

Burn marks in center of die, adjacent but
not directly under clip bondFigure 82 | Sample image 1; 15 V, 2.5 A, 100 ms

Figure 83 | Sample image 2; 15 V, 2.5 A,

		

100 msFailure signature of electrical overstress on power MOSFETs

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MOSFET & GaN FET Application Handbook

Figure 84 | Sample image 4; 15 V, 2.5 A, 100 ms

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2.16 Over-current EOS of BUK7L06-34ARC
Table 21: Over-current EOS
BUK7L06-34ARC
Cell structure:
Package:
Die size:
EOS condition:

9 mm hexagon
TO-220 (clip bond)
4.3 mm x 4.3 mm
120 A

Extensive damage starting from die where wirebonds meet die.
Secondary damage of reflowed solder and evenfused wires are visible.

aaa-004959

Figure 85 | Sample image 1

		

aaa-004960

Figure 86 | Sample image 2

aaa-004961

Figure 87 | Sample image 3

204

		

aaa-004962

Figure 88 | Sample image 4

2.17 Over-current EOS of PSMN7R0-30YL
Table 22: Over-current EOS
PSMN7R0-30YL
Cell structure:
Package:
Die size:
EOS condition:

2 μm stripe
LFPAK (clip bond)
2.3 mm x 1.35 mm
35 A, 35 ms

Burn marks in center of die, adjacent but notdirectly under clip bondSome evidence of die-cracking.

aaa-004963

Figure 89 | Sample image 6		

aaa-004964

Figure 90 | Sample image 7

aaa-004965

Figure 91 | Sample image 8		Failure signature of electrical overstress on power MOSFETs

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aaa-004966

Figure 92 | Sample image 9

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3 TablesTable 1. Examples of Human Body Model ESD failure signature.......................... 176
Table 2. UIS ruggedness test circuit and waveforms............................................... 177
Table 3. Examples of Unclamped Inductive Switching failure signature..... 178-179
Table 4. Examples of linear mode failure signature........................................ 180-181
Table 5. Examples of over-current failure signature....................................... 182-183
Table 6. Human body model EOS................................................................................ 183
Table 7. Human body model EOS................................................................................ 185
Table 8. Human body model EOS................................................................................ 186
Table 9. Human body model EOS................................................................................ 187
Table 10. Human body model EOS................................................................................ 188
Table 11. Unclamped inductive switching EOS........................................................... 189
Table 12. Unclamped inductive switching EOS........................................................... 190
Table 13. Unclamped inductive switching EOS........................................................... 191
Table 14. Unclamped inductive switching EOS........................................................... 192
Table 15. Unclamped inductive switching EOS........................................................... 194
Table 16. Linear mode EOS............................................................................................ 195
Table 17. Linear mode EOS............................................................................................ 197
Table 18. Linear mode EOS............................................................................................ 199
Table 19. Linear mode EOS............................................................................................ 201
Table 20. Linear mode EOS............................................................................................ 203
Table 21. Over-current EOS............................................................................................ 204
Table 22. Over-current EOS............................................................................................ 205

4 Figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.

Typical circuit for Human body Model ESD simulation............................ 175
Circuit diagram for UIS ruggedness test..................................................... 177
Waveforms obtained from UIS test............................................................. 177
Safe operating area; continuous and peak drain currents as afunction of drain-source voltage................................................................. 179
Figure 5. Example of maximum current rating from the data sheetof PSMN7R0-30YL.......................................................................................... 182
Figure 6. Sample image 4; after AI removal................................................................ 184
Figure 7. Sample image 4; after AI removal, close-up............................................... 184
Figure 8 Sample image 19; after AI removal.............................................................. 184
Figure 9. Sample image 19; after TEOS removal, close-up...................................... 184
Figure 10. Sample image 5; after AI removals.............................................................. 185
Figure 11. Sample image 5; after TEOS removal, close-up......................................... 185
Figure 12. Sample image 2; after AI removal................................................................ 186
Figure 13. Sample image 4; after AI removal................................................................ 186
Figure 14. Sample image 5; after AI removal................................................................ 186
Figure 15. Sample image 10; after AI removal.............................................................. 186
Figure 16. Device 4 after Al removal.............................................................................. 187
Figure 17. Device 7 after Al removal.............................................................................. 187
Figure 18. Device 5 after TEOS removal........................................................................ 187
Figure 19. Device 10 following decapsulation.............................................................. 187
Figure 20. Device 2 after Al, barrier and TEOS etch.................................................... 188
Figure 21. Device 3 after Al, barrier and TEOS etch.................................................... 188
Figure 22. Device 4 after Al, barrier and TEOS etch.................................................... 188
Figure 23. Sample image 1............................................................................................... 189
Figure 24. Sample image 2............................................................................................... 189
Figure 25. Sample image 3............................................................................................... 189
Figure 26. Sample image 4............................................................................................... 189
Figure 27. Sample image 41; 0.1 mH.............................................................................. 190
Figure 28. Sample image 43; 0.1 mH.............................................................................. 190
Figure 29. Sample image 51; 15 mH............................................................................... 190
Figure 30. Sample image 55; 15 mH............................................................................... 190
Figure 31. Sample image 6; 0.1 mH................................................................................ 191
Figure 32. Sample image 8; 0.1 mH................................................................................ 191
Figure 33. Sample image 18; 3.5 mH.............................................................................. 191
Figure 34. Sample image 20; 3.5 mH.............................................................................. 191
Figure 35. Device 1 upper (orange) hotspot................................................................. 192
Figure 36. Device 1 lower (orange) hotspot................................................................. 192
Figure 37. Device 4 upper (orange) hotspot................................................................. 192
Figure 38. Device 4 lower (orange) hotspot................................................................. 192
Figure 39. Device 6 (teal) hotspot................................................................................... 193
Figure 40. Device 7 (teal) hotspot.................................................................................. 193
Figure 41. Device 8 (teal) hotspot.................................................................................. 193

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Figure 42. Device 9 (teal) hotspot.................................................................................. 193
Figure 43. Device 1............................................................................................................ 194
Figure 44. Device 2............................................................................................................ 194
Figure 45. Device 3............................................................................................................ 194
Figure 46. Sample image 1: 15 V, 3 A............................................................................. 195
Figure 47. Sample image 2: 15 V, 3 A............................................................................. 195
Figure 48. Sample image 3: 15 V, 3 A............................................................................. 195
Figure 49. Sample image 4: 15 V, 3 A............................................................................. 195
Figure 50. Sample image 1: 30 V, 1.5 A.......................................................................... 196
Figure 51. Sample image 2: 30 V, 1.5 A.......................................................................... 196
Figure 52. Sample image 3: 30 V, 1.5 A.......................................................................... 196
Figure 53. Sample image 4: 30 V, 1.5 A.......................................................................... 196
Figure 54. Sample image 61; 20 V, 3.5 A, 30 ms........................................................... 197
Figure 55. Sample image 62; 20 V, 3.5 A, 30 ms........................................................... 197
Figure 56. Sample image 63; 20 V, 3.5 A, 30 ms........................................................... 197
Figure 57. Sample image 64; 20 V, 3.5 A, 30 ms........................................................... 197
Figure 58. Sample image 66; 20 V, 3 A, 60 ms.............................................................. 198
Figure 59. Sample image 67; 20 V, 3 A, 60 ms.............................................................. 198
Figure 60. Sample image 68; 20 V, 3 A, 60 ms.............................................................. 198
Figure 61. Sample image 69; 20 V, 3 A, 60 ms.............................................................. 198
Figure 62. Sample image 71; 30 V, 1.4 A, 60 ms........................................................... 198
Figure 63. Sample image 72; 30 V, 1.4 A, 60 ms........................................................... 198
Figure 64. Sample image 73; 30 V, 1.4 A, 60 ms........................................................... 199
Figure 65. Sample image 74; 30 V, 1.4 A, 60 ms........................................................... 199
Figure 66. Sample image 1; 15 V, 2.5 A, 100 ms........................................................... 199
Figure 67. Sample image 2; 15 V, 2.5 A, 100 ms........................................................... 199
Figure 68. Sample image 4; 15 V, 2.5 A, 100 ms ......................................................... 200
Figure 69. Sample image 5; 15 V, 2.5 A, 100 ms........................................................... 200
Figure 70. Sample image 11; 15 V, 5 A, 1 ms................................................................. 200
Figure 71. Sample image 12; 15 V, 5 A, 1 ms................................................................. 200
Figure 72. Sample image 13; 15 V, 5 A, 1 ms................................................................. 200
Figure 73. Sample image 14; 15 V, 5 A, 1 ms ................................................................ 200
Figure 74. Device 5 upper (orange) hotspot ................................................................ 201
Figure 75. Device 6 lower (orange) hotspot ................................................................. 201
Figure 76. Device 7 upper (orange) hotspot ................................................................ 201
Figure 77. Device 8 lower (orange) hotspot ................................................................. 201
Figure 78. Device 1 (teal) hotspot .................................................................................. 202
Figure 79. Device 2 (teal) hotspot .................................................................................. 202
Figure 80. Device 3 (teal) hotspot .................................................................................. 202
Figure 81. Device 4 (teal) hotspot .................................................................................. 202
Figure 82. Sample image 1; 15 V, 2.5 A, 100 ms .......................................................... 203
Figure 83. Sample image 2; 15 V, 2.5 A, 100 ms........................................................... 203
Figure 84. Sample image 4; 15 V, 2.5 A, 100 ms........................................................... 203
Figure 85. Sample image 1............................................................................................... 204
Figure 86. Sample image 2............................................................................................... 204

Figure 87. Sample image 3............................................................................................... 204
Figure 88. Sample image 4............................................................................................... 204
Figure 89. Sample image 6............................................................................................... 205
Figure 90. Sample image 7............................................................................................... 205
Figure 91. Sample image 8............................................................................................... 205
Figure 92. Sample image 9............................................................................................... 205

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Chapter 9
LFPAK MOSFET
thermal design guide
Application Note: AN90003

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1 Introduction
This chapter is a guide to assist design engineers in understanding the power
dissipation limits of the LFPAK family of packages. The maximum power that a
MOSFET can dissipate is considered as a function of the Printed Circuit Board (PCB)
design, using some common configurations. The application notes comprises ofthree main sections. The first section gives some background on MOSFET power
loss and the thermal environment. The next two sections address separately thelow power LFPAKs (LFPAK56D and LFPAK33) and the high power LFPAKs (LFPAK56
and LFPAK88).

2 MOSFET power dissipation and the design
environment
During normal operation, MOSFETs can exhibit three kinds of power losses:• Switching losses – due to voltage and current being non-zero during the 		transition between the ON and OFF states.
• Conduction losses – when the device is fully on due to its on-state resistanceRDS(on).
• Avalanche losses – if the device breakdown voltage is exceeded and an 		avalanche event occurs.
The shape of the power can change depending on the nature of the load. A
generalized model of the total power dissipated by a MOSFET is the sum of thesethree losses, see Equation (1).
Ptot = Psw + Pcond + Pav

				

[1]

In addition to electrical requirements another challenge is often the harshenvironment that a device needs to operate in, particularly in terms of212temperature. For instance, in automotive and industrial applications it’s notuncommon to encounter high ambient temperature requirements, (from 85 °C up
to 125 °C), this limits the amount of power that a MOSFET can safely handle.
Semiconductor devices are not the only parts to consider when dealing with hightemperatures. For instance, the PCB material FR4 has a maximum operating
temperature of around 130 °C, depending on manufacturer and chemistry, this ismuch lower than the limit specified for the junction of a silicon die (175 °C).
Modern applications continue to push the limits of power MOSFETs, while searching for better and better performances. As a consequence thermals have become
one of the most important aspects of systems design. One way to address thermal
issues is to carefully choose a device with the appropriate performances andprovide a good enough path through which heat can flow freely, avoiding any
impact on the device reliability.

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2.1 Heat propagation phenomena
Heat propagates because of a temperature difference between the junction andthe outside/ambient. Propagation occurs from junction and the outside/ambient
through different material, from solids (silicon, copper, FR4, aluminum) to fluids(surrounding air or even air pockets trapped in the solder joints on the PCB). Heatfinds a path whether it’s defined by an engineer or not. This means that if the pathis not designed correctly then heat might get trapped and raise the temperature of
one or multiple mediums.
The physical phenomena by which heat can propagate are: conduction, convection
and radiation.
Conduction

Ambient (Ta)Convection
Radiation

PCB

aaa-031985

Figure 1 | Thermal propogation phenomena

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2.1.1 Conduction
Conduction is the propagation of heat in a solid medium due to a temperature
difference within it, and it is caused by the random movement of atoms andmolecules. The rate of heat flow Q is directly proportional to the cross-section area
A, temperature difference and thermal conductivity k. It is inversely proportional to
the length x of the heat path, see Equation (2).
solid medium
A
Q

T2

T1

Q

x

aaa-031976

Figure 2 | Conduction in a solid medium

T1 - T2
				
Q = k · A · ________
x 				

[2]

Thermal conductivity, measured as W/(mK), is a physical property of a material anddefines its ability to conduct heat. The higher its value the higher the rate of heattransfer, therefore the better thermal conductor. From Equation 2 it can be derived
that rate of heat flow is measured in W. Therefore, heat is measured in Joule (Ws).
Table 1 lists some materials with their thermal conductivity values. Air is a very bad
conductor of heat, thus a good thermal insulator. It’s thermal conductivity is 2,000
times lower than that of steel. Notice how thermal glue has a very low thermal
conductivity of less than 2, (generic brand). Thermal glue is not a good thermal
conductor, however its main function is to create a good fit between two surfacesthat otherwise would form trapped pockets of air, which is 100 times a better
insulator than the glue itself.

Table 1: Typical thermal conductivity values
Material
Air (not moving)
Brick
Glass
Thermal glue
Steel
Brass
Silicon214

k (W/mK)
0.024
0.6
0.8
1.78
50.2
109
130

				
k (W/mK)

Material				
Aluminium
CopperSilver
Diamond

205
385
406
1000

2.1.2 Convection
Convection is the transfer of heat from a solid body to a fluid due to its movementwith respect to the surface of the body, and it is promoted by a difference intemperature between the two mediums. The fluid may be a gas (air) or a liquid.
Here the rate of heat flow depends only on surface A, temperature difference andconvection coefficient h, see Equation (3).
Tamb

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air flowTsA

aaa-031977

Figure 3 | Convection in free air

Q = h · A · (Ts - Tamb)

					

[3]

The convection coefficient is not a physical property of the fluid (like theconduction coefficient) but an abstract quantity verified by experimentation. It
depends on fluid density, velocity, viscosity, turbulence and on the solid medium
surface geometry.
The convection coefficient is measured in W/(m2K). Table 2 lists some values
measured in different scenarios of free/natural and forced cooling.

Table 2: Thermal convection coefficient values
Flow type
Gases
Air
Liquids
Phase changeh (W/m2K)
Free air2 - 2010
50 - 1,000
2,500 - 100,000

Forced cooling
25 - 300
100
100 - 40,000

As can be seen during a phase change the convection coefficient rises dramaticallyup to 100,000. This is due to the fact that during a phase change all the energy

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involved, and transferred to the fluid, is used to rearrange molecules structure anddoes not result in a temperature change.

2.1.3 Radiation
Radiation is the propagation of heat via infrared radiation.
The main benefit of radiation is that as the ambient temperature increases, and thecomponent temperature with it, the heat transfer by radiation increases as well. As
you can see in Equation (4), it depends on the fourth power of temperature. The
radiation is bigger but not by a lot and the overall effect is that radiation does nothelp a body get cooler.
T1

T2

Q

A

aaa-031978

Figure 4 | Radiation

Q = ε · σ · A · (T14 - T24)

				

[4]

The amount of radiation is determined by the surface emissivity. Emissivity of amaterial is measured between 0 and 1. A perfect emitter is called a black body
because it emits 100% of the energy it absorbs, and is assigned an emissivity valueof 1. Table 3 below gives emissivity values for different materials.

Table 3. Typical emissivity values
Material
Aluminium, polished
Aluminium, oxidized
Black electrical tape
Copper, polished
Copper, oxidized
Steel, oxidized
Water

Emissivity coefficient (ε)
0.05
0.25
0.95
0.01
0.65
0.88
0.98

A new piece of polished aluminum has emissivity of 0.05, (not good), its emissivity
increases as it oxides by 5 times. Every object emits thermal radiation, the amount216

of radiation that a particular object emits as a function of wavelength looks as abell shaped curve. Energy is emitted at all frequencies but the major part of theemission occurs at a certain wavelength range which depends on the source
temperature. The higher the temperature the higher the frequency (lower the
wavelength), that is why we see materials change color as they heat up.
At the receiving side, for most of the surfaces, the same graph looks quite flat but
to a very small range of wavelengths at which the object absorbs all the impinging
radiation. These surfaces are called selective surfaces, because they absorb only
certain wavelengths.

2.2 Thermal – Electrical analogy
When considering thermal propagation, classical methods of analysis may be used.
These are based on the thermal equations and thermal networks. These describe
the paths through which heat propagates through mediums or from one medium
to another.

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Any thermal network can be modelled by means of an electrical circuit. An analogy
for every thermal parameter can be found in the electrical domain. The respective
analogues of electric potential and current are temperature difference and rate ofheat flow. Based on these it may be observed that the thermal resistance is theratio of temperature and rate of heat flow similarly to how the voltage and current
ratio defines the electrical resistance using Ohm’s law. The main analogies may beseen in Table 4 below. Based on these, thermal networks can be solved using manyof the electrical theory laws such as Ohm’s and Kirchhoff’s laws. Also circuits can be
simplified by means of series and parallel resistor equivalences.

Table 4. Thermal and electrical analogous parameters
ThermalTemperature T (°C)
Rate of heat flow Q (W)
Thermal resistance Rth (K/W)
Thermal capacitance Cth [W·s/K]

Electrical
Voltage V (V)Current I (A)Resistance R (Ω)
Capacitance C (A·s/V = F)

An example of a thermal circuit modelled in the electrical domain using a SPICE
software may be seen in Figure 5. Figure 6 shows the response of this example
circuit as a temperature plot, where the transient and steady state thermal
behaviour may be seen.

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T2

Tfinal

T2
C2

R2

I

Tamb
Tamb

time

aaa-031975

Figure 5 | SPICE model			aaa-031984

Figure 6 | Response

2.2.1 Thermal resistance
Thermal resistance is a measure of the inertia of a material or medium towards
heat flow, just like the electrical resistance is to the movement of electrons. It istherefore a physical property of the specific component. It is calculated as the ratioof the temperature difference between two points and the rate of heat flow,
therefore as K/W. The thermal and electrical models may be seen in the pictures
below.

Thermal model

Electrical model
V1

T1
Q

T2

T - T2R th = 1QIR=

V1 - V2I

V2RCOND =L
k×A

RCONV =1
h×A

RRAD =

1
h R×A
aaa-031986

Figure 7 | Thermal and electrical resistance models

Every phenomenon governing how heat flows, namely conduction, convection andradiation, has its own thermal resistance. Each one of them depends inversely on acoefficient and surface or cross section area of the material from where heat is
generated or simply passing through. In conduction the resistance depends directly
to the length of the medium.

2182.2.2 Thermal capacitance
Thermal capacitance, sometimes known also as thermal mass, is a property of amaterial which represents how much heat (thermal energy) it can store in time,
similar to its electrical counterpart with electrical energy. Thermal capacitanceprovides also a quantity of the inertia against temperature fluctuations, the higher
the value the harder it will be to drive the stored energy in or out.

Thermal model
Q

Q×t
ΔT =
C th

Electrical modelI
ΔV =Cth = m × Cp

I×t
C

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aaa-031987

Figure 8 | Thermal and electrical capacitance models

Thermal capacitance is typically referred to using the symbol Cth, and it’s measured
as J/K. For a body of uniform composition, it can be approximated as the product ofmass (m) of the body and specific heat capacity (Cp), which is the heat capacity of asample of the substance divided by the mass of the sample.

2.2.3 Transient and steady state thermal behavior• Thermal transients describe temperatures which are changing, even at the endof the analysis window. An example may be observed below in the yellow part of
the junction temperature plot.
• Steady state thermals describe the stable region where temperatures show
minimal to no change at all. These temperature remain unchanged with thepassing of time and thus represent the final values a system might show underconstant conditions.

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Tj

Transient region - Tj is still changing

Steady-state region - Tj is constant

PD

t

t
aaa-031979

Figure 9 | Transient and steady state thermal behavior

2.3 MOSFET steady state thermal
characteristics
This section focuses on two important MOSFET thermal aspects. The junction tomounting base thermal resistance as well as the junction to ambient thermal
resistance. The aim in this chapter is to overview the two thermal paths and toshow the thermal network which describes these paths.

2.3.1 Thermal resistance junction to mounting base - Rth(j-mb)
The thermal heat path between the MOSFET junction to its mounting base is oneof the most important thermal specifications on a data sheet. It is represented by
the overall junction to mounting base thermal resistance which is a value describing
the ease with which heat is conducted along this path due to the different
materials is going through. In Figure 10, a MOSFET depiction is represented withthe thermal resistive network between the junction to mounting base.
As the MOSFET and its parts are 3 dimensional so is the mentioned heat path, thus
the 2 dimensional resistor network seen below is actually 3 dimensional in reality.
Referring back to the Rth(j-mb), it is important to remember that the value
encompasses the whole network. Lastly, the parameter describes the steady state
thermal characteristics, thus one should use average power dissipation values inorder to obtain the temperatures reached by the MOSFET.

220

Tc

Ti

Tmb

aaa-031980

Figure 10 | Thermal path junction to mounting base

2.3.2 Thermal resistance junction to ambient - Rth(j-a)
The thermal heat path between the MOSFET junction and the ambient
encompasses paths within the MOSFET itself as well as additional ones when a
MOSFET is mounted onto a PCB. Hence, heat may spread from the junction
towards the mounting base and the case. Afterwards, when the extremities of the
MOSFET are reached, the heat will flow into the surroundings via the PCB or
directly from the case. These heat paths are depicted in Figure 11.

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MOSFET & GaN FET Application Handbook

Figure 11 a) shows the heat moving from the junction to the mounting base and tothe case top and afterwards to the ambient. Figure 11 b) shows the heat movingfrom the mounting base to the PCB and through it in order to reach the ambient.

a)b)aaa-031981

aaa-031982

Figure 11 | Thermal path junction to ambient

A simplified thermal circuit may be seen in Fig. 12. It is important to notice that thetwo thermal paths from the component to the ambient are in parallel, thus
improving both may be redundant.

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Tcomponent
QPCB

Qcase

R PCB

R case-air

Tambaaa-031988

Figure 12 | Simplified Rth(j-a) thermal paths

A more cost effective method of reducing the thermal resistance between the
MOSFET to the surroundings may be to focus on one of the thermal paths andimprove it in the best way possible.
Figure 13 shows a thermal circuit which encompasses the MOSFET the PCB and theenvironment, it may be observed that the three heat spreading methods are shownby individual thermal resistors. In this particular case, the heat spreads:• Through conduction from the junction towards the outside of the MOSFET,
represented by the mounting base and the case.
• From the case to the ambient the heat propagates through convection andradiation.
• From the mounting base to the PCB and into it, conduction is again the main way
of propagation.
• From the PCB into the environment convection and conduction are the main
methods through which heat propagates to the ambient, radiation is usually
negligible.
DevicePCB

MOSFET

RCONV
TC

Tmb

TJ
RCOND

RCOND

RCOND

RCOND

RCOND

RCOND

RRAD

Junction
to mounting
base

RCONV
RRAD

RCase-Air

RCONV
RRAD

RBoard

RCONV
RRAD

RCONV
RRAD

Tamb

aaa-031983

Figure 13 | Thermal circuit, MOSFET, PCB and environment

The thermal circuit seen in Figure 13 is also represented in Figure 1. Finally, the twothermal resistor networks highlighted in green are also represented in Figure 12.

222

2.3.3 PCB thermal limitation and the 1 Watt rule
As previously described, the junction to ambient thermal path contains the board
or PCB. This is often made from FR4 material, which from a thermal and electricalperspective is an insulator. Moreover, planes, pours and traces of copper are alsopresent as these provide the circuit interconnections. Altogether we can consider
them as forming the board and giving it a certain thermal characteristic called theboard thermal resistance. It was found that this value is approximately 50-60 K/W,
depending on the amount of copper, insulation layer thickness and other factors.
Since this is specific to the materials and dimensions of a board, this limit is
imposed.
Moreover, the ambient temperature in which a board operates, the thermal
limitations of a system, the FR4 temperature limit of 120 °C to 140 °C, as well as thetemperature ranges within which a board needs to function in, gives rise to a power
limitation. These same dependencies can be noted from the following equations:

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[5]

[6]

[7]

PCB thickness, area and thermal resistivity determine the PCB thermal resistance,
given in K/W. Hence, given a specific ambient temperature and a maximum FR4
temperature, or the system maximum operation temperature, a ΔT is obtained.
Example: calculate the power dissipation of a MOSFET mounted on an FR4 PCB
within an automotive environment where the ambient temperature is 80 °C.
• The FR4 PCB material has a thermal resistance Rth(FR4 PCB) = 50 K/W• The FR4 PCB material maximum temperature (TFR4(max)) = 130 °C• The ambient temperature Tamb = 80 °C
								[8]								[9]Given the above ambient temperature and FR4 PCB characteristics, approximately
1 watt of power may be dissipated in a MOSFET within this automotive
environment.

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2.3.4 Thermal nomenclature
Terminology surrounding thermal characterization of power MOSFETs has beenrevised multiple times over the years. Regardless of these efforts, standards
organizations and semiconductor manufacturers may still use different names
when referring to the same thermal parameter, or to slightly different variations of
it. The terms often used to indicate one or the other type of thermal resistance are:
R and θ. In the case of junction to ambient, the Rth(j-amb) = Rθ(j-amb) = θ(j-amb).

3 LFPAK56D and LFPAK33
3.1 Simple configuration with a single layer
In this section, we will present the maximum power dissipation results for a simple
PCB configuration using a single layer with varying copper area.
Results for the LFPAK56D and LFPAK33 packages are given, for the LFPAK56D with
only one or with both MOSFETs conducting.

3.1.1 LFPAK56D
Set-up:• 1 layer on the top• MOSFET power dissipation is 0.1 W, 0.5 W and 1 W applied to each MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer, see Figure
14.
In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of the top copper area.

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Square of 100 mm

Length x

FR4Length x

Figure 14 | Copper area configuration: LFPAK56D, single top copper layer

The graph in Figure 15 captures two important factors:• Tj depends greatly on length “x” and thus copper area, the bigger the area thebetter the thermal performance• However, the ability of the top copper to provide heatsinking for the MOSFET
shows a “law of diminishing returns”. In other words, we cannot keep on adding
more copper area in the hope of continuing to reduce Tj. As can be seen from
Figure 15 below, Tj will plateau at around 50 °C (for 0.5 W per MOSFET) no
matter how much copper area we provide.
Care must be taken for Tj above 120 °C as PCB temperature directly under thetransistor would be close to the MOSFET Tj.
Tj
(°C)

180

(1) P = 2 x 0.1 W

160(2) P = 2 x 0.5 W(3) P = 2 x 1 W

140(4) PCB limit

(4)

120
100

(3)

80

(2)

60
40

(1)

20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 15 | Junction temperature as a function of copper side length x for LFPAK56D

225

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An alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed
is shown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB.
In Figure 16 the maximum power for the conditions given is as follows:
Tamb = 20 °C: Max power is 2.6 W per MOSFET (2 × 2.6 W permissible in thispackage)
Tamb = 80 °C: Max power is 1.65 W per MOSFET (2 × 1.65 W permissible in thispackage)
3
Pmax
(W)
2.5

(1) Power per MOSFET, Tamb = 80 °C(2) Power per MOSFET, Tamb = 20 °C

2

(2)

1.5
(1)
1
0.5

0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 16 | Maximum permissible power dissipation as a function of copper side length x for
LFPAK56D

3.1.2 LFPAK56D only one MOSFET active at a time
In a typical half bridge application only one MOSFET conducts at a time. In thegraph below, Figure. 17, 1 W is dissipated in the left MOSFET (blue curve). We cansee that 1 W dissipation in one MOSFET is not equivalent to 0.5 W dissipation ineach of the two MOSFETs (yellow curve).

226

180

Tj
(°C)

(1) P = 0.5 W

160(2) P = 1 W, left side(3) P = 2 x 1 W

140(4) PCB limit

(4)

120
100

(3)
(2)

80

(1)

60
40
20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 17 | Junction temperature as a function of copper side length x for LFPAK56D; one MOSFET
conducting

As can be seen temperature is higher in the case of one MOSFET dissipating 1 W
than it is with two MOSFETs each dissipating 0.5 W. When only one MOSFET is
active the second MOSFET does not make a significant contribution to the total
dissipation capability - meaning that if one MOSFET is off the heating is not sharedequally between the two. This is further explained by the thermal network shownin Figure 19.

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

An alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed
is shown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB. In Figure 18 the maximum power for the conditions given is as follows:• Tamb = 20 °C: Max power is 3.35 W with only left MOSFET• Tamb = 80 °C: Max power is 2.1 W with only left MOSFET
4(1) Power per MOSFET, Tamb = 80 °C

Pmax(2) Power per MOSFET, Tamb = 20 °C

(W)

3

(2)

2

(1)

1

0

0

10

20

30

40

50

Layer top side only, length "x" (mm)Figure 18 | Maximum power as a function of side length x for LFPAK56D; one MOSFET conducting

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The concept of the second MOSFET half-sharing the thermal dissipation whenturned off is not true
– see Figure 19
Dual MOSFET thermal resistance configuration:
We can see that the thermal path between both MOSFETs inside the package ishighly resistive (100 K/W).
Tj2

Tj1
Rth j1 to j2
100 K/W

Rth j1 to mb1
2.5 K/W

Rth j2 to mb2
2.5 K/W
Rth mb1 to mb2
50 K/W

Tmb1

Tmb2

Rth mb2 to amb

Rth mb1 to amb
30 K/W

30 K/W

Tamb

Tamb

Figure 19 | Thermal resistance configuration

3.1.3 LFPAK33
Set-up:• 1 layer on the top• MOSFET power dissipation is 0.1 W, 0.5 W and 1 W applied in the MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer

228• There is no forced air cooling applied i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer. In thisanalysis, we will examine the variation in device junction temperature (Tj) as afunction of top copper area.
Square of 100 mm

Length x

Length x

FR4

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

Figure 20 | Copper area configuration: LFPAK33, single top copper layer

The graph in Figure 21 captures two important factors:• Tj depends greatly on length “x” and thus copper area, the bigger the area thebetter the thermal performance• However, the ability of the top copper to provide heatsinking for the MOSFET
shows a “law of diminishing returns”. In other words, we cannot keep on adding
more copper area in the hope of continuing to reduce Tj. As can be seen from
Figure 21 below, Tj will plateau at around 40 °C (for 0.5 W per MOSFET) no
matter how much copper area we provide.
Care must be taken for Tj above 120 °C as PCB temperature directly under thetransistor would be close to the MOSFET Tj.

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Tj
(°C)

180

(1) P = 0.1 W

160(2) P = 0.5 W(3) P = 1 W

140(4) PCB limit

(4)

120
100
80

(3)

60

(2)

40

(1)

20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 21 | Junction temperature as a function of copper side length x for LFPAK33

An alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed
is shown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB.
In the graph below, Figure 22, the maximum power for the conditions given is asfollows:
Tamb = 20 °C: Max power is 3.7 W in the MOSFET
Tamb = 80 °C: Max power is 2.4 W in the MOSFET4
Pmax(W) 3.5(1) Power, Tamb = 80 °C(2) Power, Tamb = 20 °C

(2)

3
2.5

(1)

2
1.5
1
0.5
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 22 | Maximum permissible power dissipation as a function of copper side length x for LFPAK33

230

3.2 Usual configuration: 4 layers + vias
In this section, we will present the maximum power dissipation results for a PCB
configuration using 4 layers + vias for dissipation on the bottom layer, with varying
copper area.
Results for the LFPAK56D and LFPAK33 packages are given, for the LFPAK56D with
only one or with both MOSFETs conducting.

3.2.1 LFPAK56D
Set-up:• 4 layers + vias (vias number increases with the copper area with a maximum of
25 vias for each side)• MOSFET power dissipation is 0.1 W, 0.5 W and 1 W applied to each MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness on all layers (external and internal) is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied i.e. only natural convection is modeled

9
LFPAK MOSFET thermal design guide

MOSFET & GaN FET Application Handbook

The common PCB stack-up is 4 layers with vias under MOSFETs to create adissipation path to the heatsink.
In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of top copper area. In Figure 23 below, we can see the vias configuration:

231

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Square of 100 mm

Length x

vias
Length x

FR4

Figure 23 | Copper area configuration: LFPAK56D, 4 layers with vias

Vias configuration: square vias used for ease of simulation.

Figure 24 | Sectional view: LFPAK56D

Table 5: Limitation of number of vias

232

X (mm)

Vias configuration

Comments

minimalfootprint

2+2 vias

6
8
10
15
20
25
30
35
40
50

9+9 vias
12+12 vias
20+20 vias
25+25 vias
25+25 vias
25+25 vias
25+25 vias
25+25 vias
25+25 vias
25+25 vias

Maximum numberof vias able to beinserted in thecopper surface

Vias information

Vias pitch 2.5 mm
Vias side length 0.7 mm
Copper thickness 70 μm
No solder fill
25 viasmaximum

The graph in Figure 25 captures two important factors:• Tj depends greatly on length “x” and thus copper area, the bigger the area thebetter the thermal performance• However, the ability of the top copper to provide heatsinking for the MOSFET
shows a “law of diminishing returns”. In other words, we cannot keep on adding
more copper area in the hope of continuing to reduce Tj. As can be seen fromthe graph in Figure 25, Tj will plateau at around 36 °C (for 0.5 W per MOSFET) no
matter how much copper area we provide.
Care must be taken for Tj above 120 °C as the PCB temperature directly under thetransistor would be close to the MOSFET Tj.
180
Tj
160
(°C)

(1) P = 2 x 0.1 W(2) P = 2 x 0.5 W(3) P = 2 x 1 W

140(4) PCB limit

(4)

120

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

100
(3)

80
60

(2)

40

(1)

20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 25 | Junction temperature as a function of copper side length x for LFPAK56D

An alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed
is shown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB.
In the graph below, Figure 26 the maximum power for the conditions given is asfollows:
Tamb = 20 °C: Max power is 5.3 W per MOSFET ( 2 × 5.3 W permissible in thispackage)
Tamb = 80 °C: Max power is 3.55 W per MOSFET ( 2 × 3.55 W permissible in thispackage)233

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Pmax
(W)

6

(1) Power per MOSFET, Tamb = 80 °C(2) Power per MOSFET, Tamb = 20 °C

5

(2)

4
3

(1)

2
1
0
0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 26 | Maximum permissible power dissipation as a function of copper side length x for
LFPAK56D

3.2.2 LFPAK56D only one MOSFET active at a time
In a typical half bridge application only one MOSFET conducts at a time. Figure 27
below shows the results for 1 W applied to the left MOSFET (blue curve). We cansee that 1 W dissipated in one MOSFET is not equivalent to 0.5 W dissipation ineach of the two MOSFETs (yellow curve).
As can be seen temperature is higher in the case of one MOSFET conducting with 1
W than it is with two MOSFETs each dissipating 0.5 W.
Tj
(°C)

180

(1) P = 2 x 0.5 W

160(2) P = 1 W, left side(3) P = 2 x 1W

140(4) PCB limit

(4)

120
100

(3)

80
60
40

(2)

20
0

(1)
0

10

20

30

40

50
60
Layer top side only, length "x" (mm)Figure 27 | Junction temperature as a function of copper side length x for LFPAK56D; one MOSFETAn alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed isshown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB.

234

In Figure 28 the maximum power for the conditions given is as follows:• Tamb = 20 °C: Max power is 6 W with only left MOSFET• Tamb = 80 °C: Max power is 4 W with only left MOSFET
7
Pmax
(W) 6

(1) Power per MOSFET, Tamb = 80 °C(2) Power per MOSFET, Tamb = 20 °C

5
(2)

4
3

(1)

2
1
0
0

10

20

30

40

50

60

Layer top side only, length "x" (mm)LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

Figure 28 | Power dissipation as a function of copper side length x for LFPAK56D; one MOSFETAs previously mentioned, when only one MOSFET is active the second MOSFET
does not make a significant contributution to the total dissipation capabilty– see Figure 19.

3.2.3 LFPAK33
Set-up:• 4 layers + vias (vias number increases with the copper area with a maximum of
25 vias for each side)• MOSFET power dissipation is 0.1 W, 0.5 W and 1 W applied in the MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness on all layers (external and internal) is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied i.e. only natural convection is modeled
The common PCB stack-up is 4 layers with vias under the MOSFETs to create adissipation path to the heatsink.
In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of top copper area.
In Figure 29 below, we can see the vias configuration:

235

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Square of 100 mm

Length x

vias
Length x

FR4

Figure 29 | Copper area configuration: LFPAK33, 4 layers with vias

Vias configuration: square vias used for ease of simulation.

Figure 30 | Sectional view: LFPAK33

Table 6: Vias configuration
X (mm)
minimalfootprint
6
8
10
15
20
25
30
35
40
50

236

Vias configuration

Comments

1 via

Maximum numberof vias able to beinserted in thecopper surface

6 vias9 vias
20 vias25 vias25 vias25 vias25 vias25 vias25 vias25 viasVias information

Vias pitch 2.5 mm
Vias side length 0.7 mm
Copper thickness 70 μm
No solder fill
25 vias maximum

The graph Figure 31 captures two important factors:• Tj depends greatly on length “x” and thus copper area, the bigger the area thebetter the thermal performance• However, the ability of the top copper to provide heatsinking for the MOSFET
shows a “law of diminishing returns”. In other words, we cannot keep on adding
more copper area in the hope of continuing to reduce Tj. As can be seen from
Figure 31 below Tj will plateau at around 33 °C (for 0.5 W per MOSFET) no matter
how much copper area we provide.
Care must be taken for Tj above 120 °C as PCB temperature directly under thetransistor would be close to the MOSFET Tj.
Tj
(°C)

180

(1) P = 0.1 W

160(2) P = 0.5 W(3) P = 1 W

140(4) PCB limit

(4)

120

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

100
80

(3)

60

(2)

40

(1)

20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 31 | Junction temperature as a function of copper side length x for LFPAK33

An alternative to the previous approach is to look at the maximum power allowed
before reaching Tj(max) = 175 °C (MOSFET absolute max). Maximum power allowed
is shown for different ambient temperature and copper length. In this example the
PCB temperature was not considered so care must be taken for the resulting heaton the PCB.
In the graph below the maximum power for the conditions given is as follows:
Tamb = 20 °C: Max power is 6.5 W in the MOSFET
Tamb = 80 °C: Max power is 4.3 W in the MOSFET

237

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Pmax
(W)

7

(1) Power per MOSFET, Tamb = 80 °C(2) Power per MOSFET, Tamb = 20 °C

6
5

(2)

4
3

(1)

2
1
0
0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 32 | Maximum permissible power dissipation as a function of copper side length x for
LFPAK33

3.3 Placement advice for improved dissipation
In this section, we will present some results for two MOSFETs placed close to eachother on a single layer PCB with varying copper area.

3.3.1. LFPAK56D
Simulation of two MOSFETs placed next to each other is carried out and checkedagainst results seen in section 2.1. The aim is to understand the dissipation effectthat the two LFPAK56D have on one another.
Set-up• 1 layer on the top side
• MOSFET power dissipation is 0.5 W applied to each MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 200 x 150 mm• Maximum PCB operating temperature of 120 °C• Copper thickness on all layers (external and internal) is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer. In thisanalysis, we will examine the variation in device junction temperature (Tj) as afunction of top copper area.

238

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MOSFET & GaN FET Application Handbook

LFPAK MOSFET thermal design guide• 2 mm gap between right and left copper layers• Simulation carried out for different length “x”• 0.5 W applied on each internal MOSFET

200 mm2 mm between layers
kept constant
Length x

Length x

150 mmFR4

Figure 33 | Copper area configuration: LFPAK56D, single top copper layer

The graph in Figure 34 shows:• The results (in green) are similar to the ones observed in section 2.1 (slightly
higher +3 °C)• This is due to the low conductivity of FR4, despite only 2 mm gap it showed noheat transfer from one copper area to the other
Tj
(°C)

180

(1) P = 0.5 W, single LFPAK56D

160(2) P = 0.5 W, two LFPAK56D next to each other(3) PCB limit

140

(3)

120

(2)

100
80
(1)

60
40
20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 34 | Junction temperature as a function of copper side length x for 2 LFPAK56D

3.3.2 LFPAK33
Simulation of two MOSFETs placed next to each other is carried out and checked

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against results seen in section 2.3. The aim is to understand the dissipation effectthat the two MOSFETs have on one another.
Set-up:• 1 layer on the top• MOSFET power dissipation is 0.5 W applied in the MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 200 x 150 mm• Maximum PCB operating temperature of 120 °C• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer. In thisanalysis, we will examine the variation in device junction temperature (Tj) as afunction of top copper area.
• 2 mm gap between right and left copper layers• Simulation carried out for different length “x”• 0.5 W applied on each MOSFET

200 mmLength x

Length x

150 mm2 mm between layers
kept constant

FR4

Figure 35 | Copper area configuration: LFPAK33, single top copper layer

The graph in Figure 37 shows:• The results for x > 20 mm are similar to the ones observed in section 2.3 (slightly
higher +3 °C)• This is due to the low conductivity of FR4, despite only 2 mm gap it showed no240

heat transfer from one copper area to the other• For x < 20 mm results show up to 20 °C higher compared to the results fromsection 2.3.
• This is due to the MOSFETs being brought closer to each other as a result of
reduced copper area – note that in this case the space between MOSFETs is half
the space between MOSFETs in the case of LFPAK56DIn Figure 36 you can see that for x = 10 mm, the distance between the LFPAK33
MOSFETs is approximatively 10 mm. Less than 20 mm apart, the MOSFETs are closeenough to heat each other, hence we start to see a temperature difference.

x = 10 mm

2 mm gap

2x + 2 = 22 mm

x = 10 mm

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook

x = 10 mm

x = 10 mm

x + 2 = 12 mm

2 mm gap

Figure 36 | Copper area configuration: 2 x LFPAK33, 2 x LFPAK56D; separation between MOSFETs

Tj
(°C)

180

(1) P = 0.5 W, single LFPAK33(2) P = 0.5 W, 2 x LFPAK33 next to each other(3) PCB limit

160
140

(3)

120

(2)

100
80
(1)

60
40
20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 37 | Junction temperature as a function of copper side length x for 2 LFPAK33

241

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3.4 Comparison between two LFPAK56 andone LFPAK56D, then one LFPAK56D and
two LFPAK33
In this section we will present some comparative results for different package
devices on a single layer board with varying copper area.

3.4.1 Two LFPAK56 to LFPAK56D
In this section the results of two single LFPAK56 MOSFETs are compared to theresults of one dual LFPAK56D MOSFET (see Section 3.1.1)
The aim is to highlight the benefit of using one LFPAK56D dual MOSFET instead oftwo single LFPAK56 MOSFETs.
Set-up:• 1 layer on the top• MOSFET power dissipation is 0.5 W applied to each MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer. In thisanalysis, we will examine the variation in device junction temperature (Tj) as afunction of top copper area.
• Same gap between copper layer was used for both single LFPAK56 and dual
LFPAK56D• Simulation carried out for different length “x”.
• 0.5 W applied on each MOSFET

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Square of 100 mm
Length x

Length x

LFPAK56D

Length x

Length x

2 x LFPAK56
FR4

Figure 38 | Copper area configuration: 1 x LFPAK56D, 2 x LFPAK56, single top copper layer

The graph in Figure 39 shows:• Overall two single LFPAK56 show better heat dissipation than one dual 		LFPAK56D by up to approximately 10 °C. This is due to the larger surface area of
the LFPAK56 drain tab giving improved heat spreading and thermal dissipation.
• Note that the 10 °C is the relative figure between the two packages, the mostimportant factor is the operating junction temperature• If there is enough margin before reaching 175 °C at the junction, then LFPAK56D
offers an attractive option due to its space saving

Tj
(°C)

180

(1) P = 2 x 0.5 W, single LFPAK56D(2) P = 2 x 0.5 W, two LFPAK56 next to each other160(3) PCB limit

140

(3)

120
100

(1)

80
60

(2)

40
20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 39 | Junction temperature as a function of copper side length x for 2 LFPAK56 and 1
LFPAK56D243

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3.4.2 LFPAK56D to two LFPAK33
In this section the results of one dual LFPAK56D MOSFET are compared to theresults of two single LFPAK33 MOSFETs (see section 4.2)
Aim is to highlight the benefit of using one LFPAK56D dual instead of two single
LFPAK33.
Set-up:• 1 layer on the top• MOSFET power dissipation is 0.5 W applied to each MOSFET• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• Maximum PCB operating temperature of 120 °C• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer. In thisanalysis, we will examine the variation in device junction temperature (Tj) as afunction of top copper area.
• Same gap between copper layer was used for both single LFPAK33 and dual
LFPAK56D• Simulation carried out for different length “x”• 0.5 W applied on each MOSFET

Square of 100 mm
Length x

Length x

LFPAK56D

FR4

Length x

2* LFPAK33

Length x

Figure 40 | Copper area configuration: 1 x LFPAK56D, 2 x LFPAK33, single top copper layer

244

The graph in Figure 41 shows:• Overall two single LFPAK33 show better heat dissipation than a dual LFPAK56D
by up to approximately 5 °C. This is due to the larger drain surface area of the
LFPAK33 offering better thermal dissipation, (less improvement than withLFPAK56 as LFPAK33 is a smaller package).
• Note that the 5 °C is the relative figure between the two packages, the mostimportant factor is the operating junction temperature.
• If there is enough margin before reaching 175 °C at the junction, then LFPAK56D
offers an attractive option due to all the advantages that one component offers
versus two in terms of PCB layout, placement, cost effectiveness, etc.
Tj
(°C)

180

(1) P = 2 x 0.5 W, single LFPAK56D(2) P = 2 x 0.5 W, two LFPAK33 next to each other(3) PCB limit

160
140

(3)

120
100
80

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(1)

60

(2)

40
20
0

0

10

20

30

40

50

60

Layer top side only, length "x" (mm)Figure 41 | Junction temperature as a function of copper side length x for 1 LFPAK56D and 2
LFPAK33

3.5 Impact of Rth(j-mb) compared to Rth(mb-a)
Dissipation losses from the MOSFET junction are not mainly limited by the thermal
resistance Rth(j-mb) as this is very low. The high thermal path for heat dissipation ispresented by the thermal resistance Rth(mb-amb) (mounting base to PCB to ambient).
Legs

Plastic
encapsulation

Junction

Leadframe
components

Rth(j-top)Die

Rth(PCB-amb)

Rth(j-mb)

PCB
Rth (PCB)

Figure 42 | View of thermal resistances in and outside the MOSFET

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Example: for the part number BUK7M15-60E (LFPAK33, 15 mΩ, 60 V) the maximum
thermal resistance junction to mounting base is 2.43 K/W:

Table 7: Thermal resistance BUK7M15-60E
SymbolParameter

Conditions Min

Typ

Max

Unit

Rth(j-mb)

thermal resistance fromjunction to mounting baseFig. 5

2.01

2.43

K/W

-

Using thermal simulation (Flotherm) with the following conditions:
0.5 W of losses in the MOSFET, air ambient is 20 °C, 35 μm copper, we can calculate
some thermal resistance.
•
		
		
•
		
•
		

Rth(j-mb) is 0.8 K/W• This is a lower value than given in the data sheet due to the simulation
using ideal conditions
As can be seen in Table 8 below, thermal resistance for other items havehigh value compared Rth j-mb
The total thermal resistance, junction to ambient, is 59.4 K/W when using
65.4 °C as (ambient) reference point.

Table 8 lists temperatures for different points captured in the simulation andshown in Figure 43.

Table 8: Breakdown of thermal resistance for a simple case

246

Thermal resistance part

Temperature (°C)

Rth (K/W)

Junction

95.1-

Mounting base

94.7

0.8

PCB under MOSFET

88.6

12.2

PCB to the right of the
MOSFET

86.6

4

Ambient air 0.5 mm over
the top of the PCB

79.1

15

Ambient air 1 mm over
the top of the PCB

65.4

27.4

AMBIENT AIR

95.1 °C
94.7 °C

65.4 °C

MOSFET
79.1 °C
PCB

88.6 °C

86.6 °C

Figure 43 | Thermal result, LFPAK33, 0.5 W, 20 °C ambient

Due to the very low thermal resistance between junction and mounting base it isvery important to take care of design surrounding the MOSFET, (i.e. thermal vias,
copper area, heat sink, water cooling, air cooling), in order to reduce the total
thermal resistance.

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4 LFPAK56 and LFPAK88
4.1 Simple configuration with a single layer
In this section, we will present the maximum power dissipation results for a simple
PCB configuration using a single layer with varying copper area.
Results are given for the LFPAK56 and LFPAK88 packages. Models used are basedon 1 mΩ LFPAK56E and LFPAK88.

4.1.1 Set-up:
•
•
•
•
•
•
•
•

1 copper layer on the top
MOSFET power dissipation is 1 W, 1.5 W and 2 W
Maximum junction temperature of 175 °C
PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm
PCB operating temperature of 120 °C and Tj of 175 °C are highlighted in graphs
Copper thickness is 2 oz./ft2 (70 μm)
The PCB is suspended in free air at ambient temperature of 20 °C
The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled
The simplest possible PCB stack-up is that of a single top copper layer, see Figure
44.

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In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of the top copper area and calculate the maximum power that can besafely dissipated in the MOSFET to reach a Tj of 175 °C.

Square of 100 mm

FR4

Length x

Length x

aaa-029865

Figure 44 | Copper area configuration: LFPAK56, single top copper layer, the configuration is thesame for LFPAK88

4.1.2 Junction temperature as a function of copper area
The graphs in Figure 45 and Figure 46 capture two important factors:• Tj depends greatly on length “x” and thus copper area, the bigger the area thebetter the thermal performance• However, the ability of the top copper to provide heatsinking for the MOSFET
shows a “law of diminishing returns”. In other words, we cannot keep on adding
more copper area in the hope of continuing to reduce Tj. As can be seen from
Fig. 45 below, for LFPAK56, 1 W profile, Tj will plateau at around 55 °C.
Note: Standard FR4 PCBs operate at a maximum temperature of 120 °C, caremust be taken for Tj > 120 °C as the PCB area directly under the transistor will beclose to the MOSFET junction temperature, (due to low Rth(j-mb)).
The graphs below also shows the absolute minimum copper area needed for Tj ≤175 °C.

248

Tj

180

(5)

(1) P = 1 W(°C) 160(2) P = 1.5 W(3) P = 2 W

140(4) PCB limit(5) Tj limit

(4)

120
100

(3)
(2)

80

(1)

60
40
20
0

0

10

20

30

40

50

60

70

Layer top side only, length "x" (mm)Figure 45 | Junction temperature as a function of copper side length x for LFPAK56

Tj

180

(5)

(1) P = 1 W(°C) 160

LFPAK MOSFET thermal design guide

9

MOSFET & GaN FET Application Handbook(2) P = 1.5 W(3) P = 2 W

140(4) PCB limit(5) Tj limit

(4)

120

(3)

100

(2)

80

(1)

60
40
20
0

0

10

20

30

40

50

60

70

Layer top side only, length "x" (mm)Figure 46 | Junction temperature as a function of copper side length x for LFPAK88

4.1.3 Maximum allowed power dissipation as a function of copper area
The maximum allowed power dissipation is shown in Figure 47 and Figure 48
below, for different ambient temperature and copper side length. In this example
the PCB temperature was not considered so care must be taken for the resulting
heat on the PCB.
From graphs in Figure 47 and Figure 48 the maximum permissible power,(Tamb = 20 °C), is 5.05 W and 5.9 W for the LFPAK56 and LFPAK88 packages
respectively:

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Table 9: Maximum power dissipation
DeviceTamb = 20 °C

Tamb = 80 °C

LFPAK56

5.05 W

3.2 W

LFPAK88

5.9 W

3.8 W

Pmax
(W)

6

(1) Tamb = 20 °C(1)(2) Tamb = 80 °C

5
4

(2)
3
2
1
0

0

10

20

30

40

50
60
70
Side length "x" of the Cu area (mm)Figure 47 | Maximum permissible power dissipation as a function of copper side length “x” for
LFPAK56

Pmax
(W)

7
(1) Tamb = 20 °C

6(1)(2) Tamb = 80 °C

5
(2)

4
3
2
1
0

0

10

20

30

40

50
60
70
Side length "x" of the Cu area (mm)Figure 48 | Maximum permissible power dissipation as a function of copper side length “x” for
LFPAK88250

4.2 Usual configuration: 4 layers + vias
In this section, we will present the maximum power dissipation results for a PCB
configuration using 4 layers + vias for dissipation on the bottom layer, with varying
copper area.
Results are given for the LFPAK56 and LFPAK88 packages.

4.2.1 Set-up:• Four layers with vias (max number of vias is 25 and for small copper areas thisnumber will decrease accordingly, see Table 10)• MOSFET power dissipation is 1 W, 1.5 W and 2 W• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• PCB operating temperature of 120 °C and Tj of 175 °C are highlighted in graphs• Copper thickness of all layers is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled

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The common PCB stack-up is 4 layers with vias under the MOSFET to create adissipation path to heatsink (no heat sink was used in simulation).
In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of copper area (same area size applied to all layers).
The configuration of the vias is shown below in Figure 49 and Figure 50:

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VIAS

PCB: FR4 / 100 mm squareaaa-029866

Figure 49 | Copper area configuration: LFPAK56, 4 layers with vias, the configuration is similar for
LFPAK88

Vias configuration: square vias used for ease of simulation.

aaa-029867

Figure 50 | Sectional view: LFPAK56

Table 10: Limitation of number of vias

252

X (mm)

Vias configuration Comments

6

9 vias8

9 vias10

20 vias15

25 vias20

25 vias25

25 vias30

25 vias35

25 vias40

25 vias45

25 vias50

25 vias60

25 viasMaximum number of vias
able to be inserted in thecopper area

25 vias maximum

Vias information

Square vias
length 0.7 mm
Vias pitch:
2.5 mm(between vias in columns)
2.0 mm(between vias in rows)
Copper thickness:
70 μm
No solder fill

4.2.2 Junction temperature as a function of copper area (4 layers andvias)
The graphs in Figure 51 and Figure 52 below show the junction temperature as afunction of drain copper area following the same trend as the single layer PCB
configuration in that:• Tj depends greatly on copper area
• The ability of the top copper to provide heatsinking for the MOSFET shows a“law of diminishing returns”.
Tj

180

(5)

(1) P = 1 W(°C) 160(2) P = 1.5 W(3) P = 2 W

140(4) PCB limit(5) Tj limit

(4)

120
100

(3)

80

LFPAK MOSFET thermal design guide

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MOSFET & GaN FET Application Handbook

(2)

60

(1)

40
20
0

0

10

20

30

40

50

60

70

Four layers with vias, length "x" (mm)Figure 51 | Junction temperature as a function of copper side length x for LFPAK56

Tj

180

(5)

(1) P = 1 W(°C) 160(2) P = 1.5 W(3) P = 2 W

140(4) PCB limit(5) Tj limit

(4)

120
100

(3)

80

(2)

60

(1)

40
20
0

0

10

20

30

40

50

60

70

Four layers with vias, length "x" (mm)Figure 52 | Junction temperature as a function of copper side length x for LFPAK88253

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4.2.3 Maximum allowed power dissipation as a function of copper area
(4 layers and vias)
From graphs in Figure 53 and Figure 54 the maximum power for a given package
and conditions are as follows:

Table 11: Maximum power dissipation
DeviceTamb = 20 °C

Tamb = 80 °C

LFPAK56

9.6 W

6.3 W

LFPAK88

10.65 W

6.9 W

12
Pmax

(1) Tamb = 20 °C(W) 10(2) Tamb = 80 °C

(1)

8
(2)

6
4
2
0

0

10

20

30

40

50
60
70
Side length "x" of the Cu area (mm)Figure 53 | Maximum permissible power dissipation as a function of copper side length “x” for
LFPAK56
12
Pmax
(W)
10

(1) Tamb = 20 °C(1)(2) Tamb = 80 °C

8

(2)

6
4
2
0

0

10

20

30

40

50
60
70
Side length "x" of the Cu area (mm)Figure 54 | Maximum permissible power dissipation as a function of copper side length “x” for
LFPAK88254

4.3 Simple configuration with a single split
layer of copper
In this section, we will present the maximum power dissipation results as per theprevious section for a simple PCB configuration using a single layer and varying
copper area, but with copper layer split in two part (one part placed under thedrain tab of the MOSFET and the other under the source pins).
Results are given for the LFPAK88 package only.

4.3.1 Set-up:• 1 copper layer on the top - split into two areas:• 3/5 of area under drain tab• 2/5 of area under source pins
• MOSFET power dissipation is 1 W, 1.5 W and 2 W• Maximum junction temperature of 175 °C• PCB material is standard FR4, 1.6 mm thickness, dimension 100 x 100 mm• PCB operating temperature of 120 °C and Tj of 175 °C are highlighted in graphs• Copper thickness is 2 oz./ft2 (70 μm)• The PCB is suspended in free air at ambient temperature of 20 °C• The simulation is carried out for conduction, convection and radiation heattransfer• There is no forced air cooling applied, i.e. only natural convection is modeled

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The simplest possible PCB stack-up is that of a single top copper layer, see Figure
55.
In this analysis, we will examine the variation in device junction temperature (Tj) as
a function of the top copper area.

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Square of 100 mm

PCB: FR4

aaa-029871

Figure 55 | Split copper area configuration: LFPAK88, single top layer with copper

The graph in Figure 56 captures what has been previously mentioned in terms ofcopper area and heat dissipation i.e. the bigger the area the better the thermal
performance. More importantly in this configuration, it shows the importance inconsidering the source pins of an LFPAK MOSFET as a thermal path for efficiently
dissipating heat.
The graph in Fig. 56 also shows that coper area of length “x” = 40 mm in split
copper configuration provides a performance equivalent of that provided in thesolid (non-split) copper area of length “x” = 60 mm.
Split copper configuration for length “x” = 40 mm is as follows:• 24 mm x 40 mm copper area placed under the drain tab of the LFPAK MOSFET• 16 mm x 40 mm copper area placed under the source pins of the LFPAK MOSFET
Note: Standard FR4 PCBs operate at maximum temperature of 120 °C, care must
be taken for Tj > 120 °C as PCB area directly under the transistor would be close
to the MOSFET junction temperature (due to low Rth(j-mb)).
Results are for LFPAK88, but the principle applies to all Nexperia clip bond LFPAK
devices. Tamb = 20 °C: Max power of ~6 W is achieved with an area of 40 mm x 40
mm single copper layer in split configuration, whist previously shown to require anarea of 60 mm x 60 mm for single solid copper layer.

256

Pmax
(W)

7

(1) Split single layer(2) Solid single layer

6

(1)

(2)

5
4
3
2
1
0

0

LFPAK88

10

20

30

40

50
60
70
Side length "x" of the Cu area (mm)Figure 56 | Maximum permissible power dissipation as a function of copper side length “x” for
LFPAK88

LFPAK MOSFET thermal design guide

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MOSFET & GaN FET Application Handbook

4.4. Impact of Rth(j-mb) compared to		 Rth(mb-amb)
The thermal resistance Rth(j-mb) of the MOSFET is very low and therefore dissipation
losses are mainly limited by the high thermal resistive path presented by Rth(mb-amb)(mounting base to ambient).
Legs

Plastic
encapsulation

Junction

Leadframe
components

Rth(j-top)Die

Rth(PCB-amb)

Rth(j-mb)

PCB
Rth (PCB)

Figure 57 | View of thermal resistances in and outside the MOSFET

Example: for the part number BUK7S1R0-40H (LFPAK88, 1 mΩ, 40 V) the maximum
thermal resistance junction to mounting base is 0.4 K/W, see Table 12:257

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Table 12: Thermal resistance BUK7S1R0-40H
SymbolParameter

Rth(j-mb)

thermal resistance fromjunction to mounting baseConditions Min
-

Typ

Max

Unit

0.35

0.4

K/W

Using thermal simulation (Flotherm) with the following conditions:• 1 W of losses in the MOSFET• Ambient air temperature of 20 °C• 70 μm copper
we can calculate the thermal resistance for different paths, example:••••

1 W of losses in the MOSFET
Junction temperature = 52.2 °C
Mounting base temperature = 52.0 °C
RƟ = ∆T / P => Rth(j-mb) = 0. 2 K/W.

This is lower than the measured value given in the data sheet due to simulation
using ideal conditions.
As can be seen in Figure 58 below, the thermal resistance between mounting baseand ambient is of much higher value (~30 K/W) than Rth(j-mb).

258

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Figure 58 | Thermal resistance LFPAK88: single layer copper profile (50 x 50 mm)

Figure 59 | Thermal resistance LFPAK88: single layer copper profile (40 x 40 mm)

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Figure 60 | Thermal resistance LFPAK88: single split layer copper profile (40 x 40 mm: split into 24 x
40 mm and 16 x 40 mm)

5 Conclusion
All the LFPAK packages offer a very good junction to mounting base thermal
performance, meaning that the mounting base can be near to the junction
temperature, but this is often limited by the PCB high temperature capability.
It is very important for designs to reduce the thermal resistance between mounting
base and the ambient environment as this will present the bottleneck in heatdissipation. All of Nexperia LFPAK packages use clip bond technology making theirsource pins a good thermal path in addition to the thermal path provided by thedrain tab. To take full advantage of this feature, it is important for PCB layout
designs to consider placing a good amount of copper under the source pins. The
drain tab still presents the main thermal path for heat dissipation and should be thefocus for any thermal design layout.
In all cases a configuration with 4 layers with vias substantially improves the heatdissipation.
This thermal guide establishes the necessary principles in thermal design
approaches, combined with LFPAK packages features (i.e. low Rth(j-mb) and source
clip bond) offer the designer good options in optimizing PCB thermal design.
Good thermal design practices should be applied to take advantage of the very
good thermal performance LFPAK packages and Tj(max) must be kept < 175 °C forsafe operation.

260

LFPAK Vs DPAK thermal
performance comparison

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6 TablesTable 1. Typical thermal conductivity values.....................................................214-215
Table 2. Thermal convection coefficient values.......................................................215
Table 3. Typical emissivity values................................................................................216
Table 4. Thermal and electrical analogous parameters..........................................217
Table 5. Limitation of number of vias.........................................................................232
Table 6. Vias configuration...........................................................................................236
Table 7. Thermal resistance BUK7M15-60E..............................................................246
Table 8. Breakdown of thermal resistance for a simple case.................................246
Table 9. Maximum power dissipation.........................................................................250
Table 10. Limitation of number of vias.........................................................................252
Table 11. Maximum power dissipation.........................................................................254
Table 12. Thermal resistance BUK7S1R0-40H.............................................................258

7 Figures
Figure 1. Thermal propogation phenomena.............................................................213
Figure 2. Conduction in a solid medium.....................................................................214
Figure 3. Convection in free air....................................................................................215
Figure 4. Radiation.........................................................................................................216
Figure 5. SPICE model....................................................................................................218
Figure 6. Response.........................................................................................................218
Figure 7. Thermal and electrical resistance models.................................................218
Figure 8. Thermal and electrical capacitance models..............................................219
Figure 9. Transient and steady state thermal behavior...........................................220
Figure 10. Thermal path junction to mounting base..................................................221
Figure 11. Thermal path junction to ambient..............................................................221
Figure 12. Simplified Rth(j-a) thermal paths...................................................................222
Figure 13. Thermal circuit, MOSFET, PCB and environment.....................................222
Figure 14. Copper area configuration: LFPAK56D, single top copper layer...........225
Figure 15. Junction temperature as a function of copper side length x for ................
		LFPAK56D........................................................................................................225
Figure 16. Maximum permissible power dissipation as a function of copper		side length x for LFPAK56D.........................................................................226
Figure 17. Junction temperature as a function of copper side length x for ................
		LFPAK56D; one MOSFET conducting.........................................................227
Figure 18. Maximum power as a function of side length x for LFPAK56D;		
one MOSFET conducting..............................................................................227
Figure 19. Thermal resistance configuration...............................................................228
Figure 20. Copper area configuration: LFPAK33, single top copper layer..............229
Figure 21. Junction temperature as a function of copper side length x
		for LFPAK33....................................................................................................230

262

Figure 22. Maximum permissible power dissipation as a function of		
copper side length x for LFPAK33..............................................................230
Figure 23. Copper area configuration: LFPAK56D, 4 layers with vias......................232
Figure 24. Sectional view: LFPAK56D............................................................................232
Figure 25. Junction temperature as a function of copper side length x for ................
LFPAK56D........................................................................................................233
		Figure 26. Maximum permissible power dissipation as a function of copper		side length x for LFPAK56D.........................................................................234
Figure 27. Junction temperature as a function of copper side length x for ................
		LFPAK56D; one MOSFET..............................................................................234
Figure 28. Power dissipation as a function of copper side length x for		
LFPAK56D; one MOSFET..............................................................................235
Figure 29. Copper area configuration: LFPAK33, 4 layers with vias.........................236
Figure 30. Sectional view: LFPAK33...............................................................................236
Figure 31. Junction temperature as a function of copper side length x for ................
LFPAK33..........................................................................................................237
		

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Figure 32. Maximum permissible power dissipation as a function of copper		side length x for LFPAK33............................................................................238
Figure 33. Copper area configuration: LFPAK56D, single top copper layer...........239
Figure 34. Junction temperature as a function of copper side length x for		
2 LFPAK56D....................................................................................................239
Figure 35. Copper area configuration: LFPAK33, single top copper layer..............240
Figure 36. Copper area configuration: 2 x LFPAK33, 2 x LFPAK56D;
		
separation between MOSFETs....................................................................241
Figure 37. Junction temperature as a function of copper side length x for		
2 LFPAK33.......................................................................................................241
Figure 38. Copper area configuration: 1 x LFPAK56D, 2 x LFPAK56,		single top copper layer.................................................................................243
Figure 39. Junction temperature as a function of copper side length x for		
2 LFPAK56 and 1 LFPAK56D........................................................................243
Figure 40. Copper area configuration: 1 x LFPAK56D, 2 x LFPAK33,
		single top copper layer.................................................................................244
Figure 41. Junction temperature as a function of copper side length x for		
1 LFPAK56D and 2 LFPAK33........................................................................245
Figure 42. View of thermal resistances in and outside the MOSFET.......................245
Figure 43. Thermal result, LFPAK33, 0.5 W, 20 °C ambient.......................................247
Figure 44. Copper area configuration: LFPAK56, single top copper layer,		the configuration is the same for LFPAK88..............................................248
Figure 45. Junction temperature as a function of copper side length x for		
LFPAK56..........................................................................................................249
Figure 46. Junction temperature as a function of copper side length x for		
LFPAK88..........................................................................................................249
Figure 47. Maximum permissible power dissipation as a function of		
copper side length “x” for LFPAK56...........................................................250

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Figure 48. Maximum permissible power dissipation as a function of		
copper side length “x” for LFPAK88...........................................................250
Figure 49. Copper area configuration: LFPAK56, 4 layers with vias, the .......................
		
configuration is similar for LFPAK88..........................................................252
Figure 50. Sectional view: LFPAK56...............................................................................252
Figure 51. Junction temperature as a function of copper side length x for		
LFPAK56..........................................................................................................253
Figure 52. Junction temperature as a function of copper side length x for		
LFPAK88..........................................................................................................253
Figure 53. Maximum permissible power dissipation as a function of		
copper side length “x” for LFPAK56...........................................................254
Figure 54. Maximum permissible power dissipation as a function of		
copper side length “x” for LFPAK88...........................................................254
Figure 55. Split copper area configuration: LFPAK88, single top layer		with copper.....................................................................................................256
Figure 56. Maximum permissible power dissipation as a function of		
copper side length “x” for LFPAK88...........................................................257
Figure 57. View of thermal resistances in and outside the MOSFET.......................257
Figure 58. Thermal resistance LFPAK88: single layer copper profile		(50 x 50 mm)...................................................................................................259
Figure 59. Thermal resistance LFPAK88: single layer copper profile		(40 x 40 mm)...................................................................................................259
Figure 60. Thermal resistance LFPAK88: single split layer copper profile		(40 x 40 mm: split into 24 x 40 mm and 16 x 40 mm)..............................260

Chapter 10
Maximum continuous
currents in Nexperia
LFPAK power MOSFETs
Application Note: AN90016

Maximum continuous currents in Nexperia LFPAK power MOSFETs

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Nexperia | A Power Design Engineer's Guide

1 Introduction
This chapter examines the factors that determine the maximum permissible
current ratings for LFPAK (copper-clip package) MOSFETs.
Modern electronics using low-voltage (<100 V) MOSFETs has seen an increase inhigh-power demand in both automotive and industrial applications. Power output
in kilowatt terms for applications such as motor drive is now a very common
requirement. Combined with the existing space constraint in modules this meansthat the need to handle more power is being passed on to the components,
particularly MOSFETs.
The current limit given in data sheets for power MOSFETs is one of the mostimportant parameters in such high-power applications where the handling of very
high currents is required.
As the MOSFET is a three terminal device – Gate, Source and Drain – current canflow through any of these terminals as IG, IS and ID respectively. Only maximum
continuous current ID (drain- source) and continuous current IS (source-drain/body
diode) will be considered. Leakage currents (IGSS, IDSS) and pulsed currents such as
IDM are not in the scope of this document
This chapter gives a comprehensive insight into the methodology in determining
the maximum continuous current rating of Nexperia power MOSFETs.
It is critical to fully understand the capabilities, the boundaries and the relevant
environmental conditions so that electronics engineers and designers select theright MOSFET for the right application - all of which will be discussed andaddressed in this document.
This chapter is specific for LFPAK and its copper clip bond technology– Nexperia's flagship package.

2 LFPAK, superior performance
LFPAK packages are compact in size, they offer much higher power density andreduced parasitic inductances compared with wire bond devices. Combined with
their copper clip bond technology, they have played an important role in MOSFETs
achieving a very high current capability.
As illustrated in Figure 1 below, LFPAK copper clip bond packages have thefollowing benefits:• Prevents localised current crowding shown in Fig 1 b) D2PAK wire bond package266• Allows for a more uniform current spread• Acts as a heat sink to the die
JDC Vol
2(A/m )

9.7768e+008
9.1661e+008
8.5554e+008
7.9447e+008
7.3340e+008
6.7233e+008
6.1126e+008
5.5019e+008
4.8912e+008
4.2805e+008
3.6698e+008
3.0592e+008
2.4485e+008
1.8387e+008
1.2271e+008
6.1638e+007
5.6895e+005

JDC Vol
2(A/m )

1.4914e+008
1.3394e+008
1.3074e+008
1.2154e+008
1.1234e+008
1.0314e+008
9.3937e+007
8.4737e+007
7.5536e+007
6.6336e+007
5.7136e+007
4.7936e+007
3.8735e+007
2.9535e+007
2.0335e+007
1.1134e+007
1.9341e+006a) LFPAK88b) D2 PAK

Note: current density contour scaling is different for LFPAK88 and D2 PAK, see accompanying colour scaling

Figure 1 | Current crowding contours for a) LFPAK88 and b) D2PAK packages

3 Key parameters that set out the boundaries
Before discussing what maximum current a MOSFET can achieve, it is important tohighlight the parameters governing the boundaries of the environment within
which the MOSFET must operate. These boundaries are mainly set by the thermal
environment, but also by the conditions that impact the MOSFET data sheet
parameters; both have a direct effect on the performance and the capabilities ofthe MOSFET.

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MOSFET & GaN FET Application Handbook

Thermal environment and thermal parameters
This section discusses the thermal parameters and explains their direct impact on
the maximum current in a MOSFET.

Temperature range and the 175 °C limit-55 °C is the lowest temperature given in the data sheet. Although normally this isassociated with storage temperature, MOSFETs' characteristics in Nexperia data
sheet are given against this value. Note - the lowest temperature associated withreal life application is usually -40 °C
25 °C (unless otherwise stated) is the reference temperature that all maximum
capabilities of MOSFETs are based upon. In Nexperia MOSFET data sheets, this isgiven as a mounting base temperature parameter – Tmb – referring to the central
point of the MOSFET drain tab. Note - other MOSFET venders commonly use Tc

267

Maximum continuous currents in Nexperia LFPAK power MOSFETs

10

Nexperia | A Power Design Engineer's Guide(case temperature), which refers to the same point, i.e. drain tab and not to theplastic part of the MOSFET.
175 °C refers to the junction, i.e. the silicon die, temperature of the MOSFET andthe parameter for this is given as Tj. All MOSFETs must operate below thistemperature – more details are given later.
The 175 °C limit explainedWith high temperatures, it is understandable to think that the plastic mould shouldbe the first cause of concern.
Historically plastic moulds have caused issues and although improvements havebeen made in this industry, if the right compound is not properly selected it can stillcause issues and leads to device failures. However, plastic mould compound alone
does not give the full story nor is it the source of setting the 175 °C limit, as will be
discussed in the next section.

4 MOSFET structureThis section briefly discusses the main components in the internal structure of
LFPAK MOSFETs and their temperature properties:

Copper (Source) clip
one piece
Silicon die

LFPAK

Plastic mould
compound

Solder attach
Copper Drain Tab

Figure 2 | LFPAK internal structure

Silicon / die: 250 °C. Pure silicon melting temperature is 1,414 °C. However, siliconin the MOSFET is doped and at ~250 ⁰C there will be thermal activation where
current will flow across the PN junction and the MOSFET no longer acts as aswitching device – i.e. there is no control over turn-on / turn-off.
Copper clip: melting temperature of copper is 900 °C. This is to highlight the fact

268

that it can handle very high temperature and not used for any calculation purposes.
The criteria used to make sure that the copper in the MOSFET is capable for current
handling is explained later in the note.
Solder attach: melting point >300 °C.
Plastic mould compound: Can potentially harden and becomes brittle around
190 °C and above. The composition of the plastic mould compound is carefully
selected to withstand high temperature specification.
As shown above, the internal components of the MOSFET are either naturally
capable of withstanding temperatures >175 °C or specifically selected to do so.
Limiting the MOSFETs maximum junction temperature Tj to 175 °C is driven by thereliability requirements MOSFETs need to meet. And thus, 175 °C is thetemperature limit used by Nexperia for qualification and life test of MOSFETs inline with industry standard.
All automotive power MOSFETs must meet the 175 °C junction temperature
specification. Although this requirement is not applicable to non-automotive
devices which meet Tj of 150 °C, most of Nexperia industrial MOSFETs are life
tested and qualified to 175 °C.

5 Maximum power and maximum current
5.1 Maximum power

10
Maximum continuous currents in Nexperia LFPAK power MOSFETs

MOSFET & GaN FET Application Handbook

With the junction temperature limit set to a maximum of 175 °C, the maximum
amount of power allowed in the MOSFET can then be determined.
The key parameters needed to calculate this maximum power allowance are thethermal impedance between the die and the mounting base; Zth(j-mb) and thermal
resistance between the die and the mounting base Rth(j-mb).
Rth(j-mb) is the thermal resistance which means that the thermal response hasreached steady state conditions (also referred to as DC conditions).
Zth(j-mb) is the term used to represent thermal impedance in its entirety, steady state
as well as transient conditions (more details are given later).

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Plastic mould
Copper clip
Source pins
Silicon die
Rth(j-top)

Solder attach
Copper Drain Tab

Rth(PCB-amb)
Rth(j-mb)

PCB
Rth(PCB)

Figure 3 | Thermal resistances inside and outside the MOSFET

Figure 3 illustrates the main thermal paths that exist from the MOSFET silicon die
to the external surroundings and ambient environment. The dominant path indissipating heat from the die is through the MOSFET drain tab and therefore thethermal impedance of most relevance is Zth(j-mb) (also referred to as Rth(j-mb)).
It is worth pointing out that the source clip in the LFPAK package also provides animportant thermal path. Having the right amount of copper in the PCB layout forthe source pins is beneficial and should be considered in the design.
The LFPAK thin plastic mould adds another option of dissipating heat from the topof the device should the design consider a heat sink at the top. For more details
about thermal performance and recommendations please refer to the applicationnote AN90003 (chapter 9 of this book) - LFPAK MOSFET thermal design.
As previously mentioned all maximum capabilities of the MOSFET are given inreference to Tmb = 25 °C.
The maximum power allowance can then be derived from the following formula:							

[1]

The junction to mounting base thermal impedance values can be obtained fromthe graph provided in the data sheet; the parameter used is transient thermal
impedance Zth(j-mb).
For references the device used in the application note is PSMNR70-40SSH(LFPAK88 0.7 mΩ, 40 V, standard level) qualified to 175 °C.

270

1
Z
Zth(j-mb)(K/W)

Steady state /DC region

Transient region
δ = 0.5

10-1

0.2
0.1
0.05
P

10-2 0.02

δ=single shot
tp
10-3
10-6

10-5

10-4

10-3

10-2

10-1

tp
T

t
T
tp (s)

1

aaa-029191

Figure 4 | Transient thermal impedance from junction to mounting base as a function of pulse
duration

As can be seen from the graph in Fig. 4 the MOSFET thermal response is similar toan RC network electrical response – hence the thermal models provided on
Nexperia support page representing this response are referred to as RC thermals– see the Support tab of the PSMNR70-40SSH product information page. For
detailed information about RC thermal models see application note AN11261(chapter 5 of this book) - Using RC Thermal models.
The curve in the graph showing pulses up to 100 ms is known as the transientcondition section (transient region) and the parameter used is Zth(j-mb). It is given insingle shot pulse or repeated PWM pulses with various duty cycles. The transientregion is relevant to situations such as short circuits, power surges or switchingtransitions, they tend to be high power for short periods of time. Larger devices
normally perform better in this region due to their bigger drain tab areas.

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MOSFET & GaN FET Application Handbook

For pulses above 10 ms, as can be seen from the graph the curves start to plateau
and will flatten after 100 ms. This section of the graph is referred to as steady stateand is given as thermal resistance parameter Rth(j-mb).
From 100 ms onwards the MOSFET will transition into thermal stability and isconsidered in DC state. In this region the thermal impedance reaches its maximum
value and the steady state capabilities of the MOSFETs are given against thismaximum value.
Note: the time it takes the MOSFET to reach steady state is not necessarily thesame for the PCB where the MOSFET is mounted on as the PCB thermal response isslower. Therefore it is important to use the right thermal impedance for each the
PCB and the MOSFET when running thermal analysis for a given condition. RC
Cauer models found in Nexperia support page allow for PCB RC network to be
added if known.

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5.2 Maximum continuous drain current
The maximum current a MOSFET can achieve is primarily derived from the
maximum power allowance in the MOSFET. When calculating maximum continuous
current, the maximum steady state power must be used.
Example:Device Name: PSMNR70-40SSH
Tmb = 25 °C
Tj(max) = 175 °C
Rth(j-mb) = (0.4 K/W max)

Table 1. Thermal characteristics
SymbolParameter

Conditions Min

Typ

Max

Unit

Rth(j-mb)

thermal resistance fromjunction to mounting base

Figure 4

0.35

0.4

K/W

-

							

[2]This value for maximum power can be found in data sheet limiting values table:

Table 2. Limiting values
In accordance with the Absolute Maximum Rating System (IEC 60134).
SymbolParameter

Conditions

Min

Max

Unit

Ptot

total power dissipation

Tmb = 25 °C

-

375

W

Using the power formulae:
P = I2× R

				

[3]

Where I is the drain current (ID) and R is the on-state resistance of the MOSFET(RDS(on)). ID can then be calculated as:
[4]

272

The on-state resistance value that must be used to calculate ID(max), is the MOSFETRDS(on) at Tj max – in this case RDS(on) at Tj = 175 °C.
[5]A factorisation graph is provided in the data sheet for RDS(on) as a function ofjunction temperature, (see Figure 5). This can be useful to calculate the current fora specific temperature requirement. For PSMNR70-40SSH the RDS(on) multiplication
factor for Tj = 175 °C is 2.19. Note: this graph is based on measured values.
a

2.4
2
1.6
1.2
0.8
0.4
0
-60

-30

0

30

60

90

RDS(on)ɑ = −RDS(on) 25 °C

120 150
Tj (°C)180

aaa-026897

Figure 5 | Normalized drain-source on-state resistance factor as a function of junction temperature

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MOSFET & GaN FET Application Handbook

From the data sheet characteristics table,
Max RDS(on) = 0.7 mΩ (VGS = 10 V, Tj = 25 °C):

Table 3. Characteristics
Symbol Parameter

Conditions

Min

Typ

Max Unit

VGS = 10 V; ID = 25 A; Tj= 25 °C

0.43 0.62 0.7

VGS = 10 V; ID = 25 A; Tj= 175 °C

0.85 1.23 1.53 mΩ

Static characteristics

RDS(on)drain-source on-state
resistancemΩThe resulting RDS(on) at VGS = 10 V, Tj = 175 °C = 1.53 mΩ, (2.19 x 0.7 mΩ). This value
is also included in the data sheet characteristic table. For further details about
MOSFET data sheet parameters please refer to application note AN11158 (chapter

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Nexperia | A Power Design Engineer's Guide

1 of this book) - Understanding power MOSFET data sheet parameters..

[6]495 A is considered to be the theoretical capability at Tj = 175 °C. Another termcommonly used is silicon capability.
Once the theoretical maximum ID is established, the next stage is to validate thisvalue through test and verification. This will allow for other limiting factors to behighlighted and considered in finalising and protecting the ID(max) given in the data
sheets.
In the case of PSMN70-40SSH, the validated ID(max) @ Tmb = 25 °C is 425 A.

Table 4. Limiting values
In accordance with the Absolute Maximum Rating System (IEC 60134).
Symbol ParameterIDdrain current

Conditions
VGS = 10 V; Tmb = 25 °C
VGS = 10 V; Tmb = 100 °C

Min
[1]

Typ

Max Unit

0.43 0.62 0.7

mΩ

-

mΩ

350

A

[1] 425A. Continuous current has been successfully demonstrated during application. Practically, thecurrent will be limited by the PCB, thermal design and operating temperature.

The formulae:

can also be used for different temperatures. For example ID(max) at 100 °C can becalculated as follows:
[7]

[8]

The same principle applies across the full operating temperature range. MOSFET
data sheets contain a drain current de-rating graph in the limiting values section,
see Figure 6.

274

500
ID
(A)
(1)

400

300

200

100

0

0

25

50

75

100

125

VGS ≥ 10 V

150 175
Tmb (°C)

200

aaa-029500(1) 425 A continuous current has been successfully demonstrated during application tests. Practicallythe current will be limited by PCB, thermal design and operating temperature.
Figure 6 | Continuous drain current as a function of mounting base temperature

Fig. 6 shows:• ID(max) for silicon capability @ 175 °C (solid curve)• ID(max) capped value (dashed line)
As stated in Nexperia data sheets the continuous ID(max) value in the limiting table is
not given as a figure verified by design i.e. theoretical, but rather demonstrated
through testing. Some of the reasons for capping the continuous ID(max) value atlower limits than the solid curve given in the graph are as follows:
Package limit

Maximum continuous currents in Nexperia LFPAK power MOSFETs

10

MOSFET & GaN FET Application Handbook• This is normally associated with wire bond MOSFETs.
• Historically old parts (10 years or older) experienced issues with the package- this was due to a combination of:
1. Wire bond fuse - being a major reason for lowering the current rating
2. Plastic mould compound.
• This limited ID(max) to 100 A - 120 A, and this legacy approach had been carried on
for some devices even if they were clip bond LFPAK and less than 10 years old.
• As MOSFETs capabilities have reached a much higher level than has been 		possible in the past, it is only appropriate to select the right plastic compoundand adapt a more accurate approach so that these new limits are achieved.
• By making the package not the limiting factor Nexperia MOSFETs can operate
to their optimum level, giving the designer full advantage of these capabilities.
Test boards• Test boards are important part in verifying the ID(max) capabilities more so at275

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Nexperia | A Power Design Engineer's Guide

Nexperia as they may be the limiting factor – as mentioned previously onlyID(max) values that are verified through test are used.
• Improvement have been made in recent years so the right test boards are used
to maintain Tmb at 25 °C.
•
Note that some MOSFETs vendors state continuous max current as		verified by design and some limit the rating to a one second test. 		
		
Although for Rth thermal stability is considered to be reached at one
		second, (represented by the flat line in Zth graph) and for the MOSFET this		
is steady state condition. The length of time for max ID test at Nexperia		
exceeds 30 seconds continuous.
Tj(max) exceeded• Junction temperature Tj is monitored during continuous ID testing. Higher ID
values resulting in Tj exceeding 175 °C are not considered.
Silicon limit• The silicon capability at 175 °C given in the data sheet as the solid curve is theabsolute limit. Meaning only the ID values that PASS the test at the curve level or
below are validated. Instances where MOSFETs ID measure values above thecurve can be explained by the fact that most MOSFETs operate at their typicalRDS(on) values while the curve only considers the RDS(on)(max) values.
Source pins
• Calculations based on the source pins dimensions i.e. length, width, cross-section
area, size, etc., as well as the alloy/copper property are used to make sure thecurrent density and capability of the pins meet the ID rating.
In summary the final maximum continuous ID rating is based on the lowest
limit met by any of the above criteria.

5.3 Maximum continuous source current
Although IS has always been given as a separate parameter, its value had historically
been linked to ID. The value of IS was based on calculation and was either lower thanID or made the same even if it was calculated to be a higher value. A more accurate
approach is to rate the current capabilities for both ID and IS separately. Thisapproach is now standard and verification through testing in the same way as
explained previously is applied.

276

Determining the IS limiting value
As far as the maximum power in the MOSFET is concerned, it is the same power
allowance whether it is applied through the MOSFET channels (ID) or the MOSFET
body diode (IS). Therefore the simple power formulae to use is P = V × I, where V isthe body diode voltage drop (VSD) and I is the source-drain current IS. Therefore thepower calculated in the previous sections still applies, and in the case of the device
example given, P = 375 W.
• The power through the MOSFET channel is P = I2 × R• Power through the MOSFET body diode is P = V × I, where V is the diode voltage
drop (VSD) and I is the source-drain current IS
The VSD maximum value given in the data sheet characteristics table is 1 V, (seeTable 5 below).

Table 5: Characteristics
Symbol Parameter

Conditions

Min

Typ

Max

Unit

0.75

1

V

Source-drain diodeVSD

source-drain voltage

IS = 25 A; VGS = 0 V; Tj= 25 °C

In theory IS should be = 375 A, (I = P/V, P = 375 W, V = 1). In reality though VSD istypically around 0.75 V and this would be the case for more than 90% of all devices.
Furthermore, VSD is temperature dependant and the voltage will drop as
temperature goes up. It is also true to say that the voltage VSD goes up whencurrent increases, but the heating element (induced by the current as self-heating)has a bigger impact.

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MOSFET & GaN FET Application Handbook

The continuous source current that has been measured and is validated in the data
sheet for PSMNR70-40SSH is 500 A, see Table 6 below.

Table 6. Limiting values
In accordance with the Absolute Maximum Rating System (IEC 60134).
Symbol Parameter

Conditions

Min

Max

Unit

-

500

A

Source-drain diodeIS

source current

Tmb = 25 °C[1][1] 500A. Continuous current has been successfully demonstrated during application. Practically, thecurrent will be limited by the PCB, thermal design and operating temperature.

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Source-drain characteristic
MOSFET data sheets include a VSD characteristic graph, see Figure 7 below.
400
IS
(A)
320

240

160
175°C
80

0

Tj = 25°C

00.2

VGS = 0 V

0.4

0.6

0.8

1
VSD (V)

1.2

aaa-029198

Figure 7 | Source-drain (diode forward) current as a function of source-drain (diode forward) voltage;
typical value

The above graph is obtained by testing the device with pulsed current to avoid
self-heating, (which would result in higher values for VSD). In real applications
self-heating will most likely occur, as currents lasting more than 100 ms will be
sufficient to get Rth in steady state region and allow the device to heat up.
To help determine VSD values at higher temperatures Figure 7 includes curves for Tj= 25 °C and for Tj = 175 °C. This shows that the VSD value reduces with increased
junction temperature. As can be seen from Figure 7 (Tj = 175 °C curve), with IS = 400
A, VSD is just over 0.8 V and with power = 375 W this will result in IS = 440 A. The
rating for IS given in data sheet is = 500 A, this difference can be explained by thefollowing: power of 375 W is based on max Rth of 0.4 K/W. However, a typical Rth is= 0.35 K/W resulting in power = 428 W. With IS = 500 A, VSD = 0.85 V.
Note: the IS value given in the data sheet (in this case 500 A) has been proventhrough testing and the value is validated.

6 Practical application examples
This section details the function and configuration of power MOSFETs in a basic
application circuit. Three key application stages will be discussed - RPP (Reverse
Polarity Protection), isolation and load drive. Figure 8 below shows a simplified
circuit capturing some basic operations of the different MOSFET configurations.

278

CableRc

Lc

Filter
Lf

S

D

Cf

Output Load drive
circuit1

G
D

S

L

R

L

Motor drive

R
ILoad

T2

T1

G

DC/Battery
SupplyIsolator

Reverse Polarity
Protection

ITotal

I1

...x3

C

C

T4

M

T5
Lf

Isolator

Output Load drive
circuit2

G
D

S

L

R

DC-to-DC converter

R

L

ILoad

T3

I2

C

T6

L

C

T7
C

aaa-031929

Figure 8 | Simplified motor drive and DC-to-DC converter application diagram

6.1 Output load drive MOSFETs T4, T5, T6
and T7The circuits shown in Figure 9 and Figure 10 both show a pair of power MOSFETs
implemented as a half-bridge, a common configuration for motor drive (T4 and T5
see Figure 9) and DC-to-DC converters (T6 and T7, see Figure 10) – for details on
half-bridge MOSFET switching parameters and performance refer to Nexperia
application note AN90011 (chapter 7 of this book) - Half-bridge MOSFET switching
and its impact on EMC.
Output Load drive
circuit1
L

R

L

R

L
I LoadC

Output Load drive
circuit2

Motor drive

C

R

L

I LoadM

CDC-to-DC converter

R

T4

...x3

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MOSFET & GaN FET Application Handbook

C

T6

L
T7

T5

C

aaa-031931

Figure 9 | Output load drive circuit 1 (motor drive)

aaa-031932

Figure 10 | Output load drive circuit 2					(DC-to-DC)279

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280

Nexperia | A Power Design Engineer's Guide

In this example we look at output load drive circuit 1, T4 and T5 in relation to
current demand which will be gated by the type of motors driven and the power
requirement of the application.
Motor drive application
Parameters profiling a motor can be summarised in Figure 11 below:
Speed
V2 > V1

V1

Volt

Max Power
ag

ea
pp

lie
d

rr
Cu

en

t

Normaloperatingarea
TorqueMax speed
No loadMax torquestall current

aaa-031933

Figure 11 | Motor operation parameters

Normal operating area is where a motor is designed and specified to spend most of
its working life. However, it is expected that motors will go through situations ofstalled rotation a number of times in their operating life. These stalled motor
situations, albeit don’t happen on a prolonged periods, are considered part of the anormal operation. Adding to this profile, fault conditions situation such as shortcircuit where motor drive circuit are expected to recover from, one can appreciate
the type of power capabilities that need to be considered in selecting a power
MOSFET driving such application. Note: although stall conditions may last few tens
or hundreds milliseconds, as explained previously this will be enough to make the
MOSFET operate in DC conditions and therefore continuous current capability is
key.

6.2 Output isolator MOSFETs T2 and T3
Isolator

Output Load drive
circuit1

G
D

S

L

R

L

ILoad

T2

I1

Motor drive

R

...x3

C

C

T4

M

T5

Isolator

Output Load drive
circuit2

G
D

S

L

R

L

ILoad

T3

I2

C

DC-to-DC converter

R

C

T6

L
T7
C

aaa-031934

Figure 12 | MOSFETs used as isolators T2 and T3

Some designs may drive two different output loads. A representation of suchcircuits is given in Figure 12 above. MOSFETs T2 and T3 are used to isolate theirrespective circuits as they may not need to operate at the same time.

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MOSFET & GaN FET Application Handbook

MOSFETs are also used as isolators is designs using the same circuit duplicated todrive one common output. This may be due to redundancy requirement in order tomeet a safety standard (usually associated with automotive application to allow
continuous operation even in cases of failure – such redundancy circuit is not partof this example here, but the same principle applies).
In such applications particular attention should be given to:• Current capability in MOSFETs. T2 and T3 need to cater not only for the current
needed to drive the loads, but also for the immediate drive circuits current
drawn (Figure 12 circuit 1 & circuit 2 respectively).

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Maximum continuous currents in Nexperia LFPAK power MOSFETs

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Nexperia | A Power Design Engineer's Guide• Some designs require large capacitors in the drive circuit. The drive circuit itself
can be treated as a capacitive load and therefore high inrush current or soft
start capabilities are important in the selection of MOSFETs – soft start mode isapplied to MOSFET’s turn on so that high inrush currents are brought undercontrol and this type of operation requires a MOSFET with strong SOA
performance. For SOA related details please see Nexperia application note
AN11158 (chapter 1 in this book) - Understanding power MOSFET data sheet
parameters.
• Another issue that might result from such large capacitors used in the drive
circuits is the current they supply in a case of a fault incident explained asfollows: In a case where T3 is off (circuit 2 is disabled) and T2 is ON, if a shortcircuit occurs in drive circuit 1, charged capacitors from drive circuit 2 cansupply very high current to drive circuit 1 through the MOSFET body diode ofT3 and through T2 (the same can occur in the opposite direction). In thisexample it is important to pay attention to IS capability in a MOSFET
MOSFETs T2 and T3 require strong capabilities in continuous ID, IS anddepending on implementation of soft start, strong SOA.

6.3 Reverse polarity protection MOSFET T1

Cable

Rc

Lc

Filter

Lf

S

D

G

DC/Battery
SupplyCf

Isolator

Reverse Polarity
Protection

G
D

S

T1

T2

ITotalI1

Lf

Isolator
G
D

S

T3

I2

aaa-031935

Figure 13 | Reverse polarity protection MOSFET T1

See Figure 13 continuous current capability is of most relevance here as the
MOSFET T1 is implemented on the supply line which allows the flow of all the282current needed for the whole module on a continuous basis.
It can also be seen that current flows through T1 from the supply regardless ofwhether the MOSFET is ON or OFF – when MOSFET is OFF it will still conduct
through its body diode. As such it is important to pay attention to IS capability.
To avoid current flowing through MOSFET body diode some designs implement
two MOSFETs back to back for reverse polarity protection, see Figure 14.
Reverse Polarity
Protection
S

D

G

T1

D

S

T2
G
aaa-031936

aaa-031935

Figure 14 | Reverse polarity protection using back-to-back MOSFETs

In a back-to-back configuration the other parameter that becomes of relevance isthe avalanche performance of the MOSFETs. A situation where a fault condition
occurs and T2 turns off as part of a protection mechanism. Due to inductances in
the cable and/or the circuit a build-up of a voltage at the drain of T2 might occur
and potentially lead to an avalanche event. For details on avalanche topic see
Nexperia application note AN10273 (chapter 7 of this book) - Half-bridge MOSFET
switching and its impact on EMC.
Multiple MOSFET parameters are highlighted in the application examples given
above, all showing the importance of, and the need to use a MOSFET that is
capable of handling high continuous current.

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7 TablesTable 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.

Thermal characteristics.................................................................................. 272
Limiting values................................................................................................ 272
Characteristics................................................................................................. 273
Limiting values................................................................................................ 274
Characteristics................................................................................................. 277
Limiting values................................................................................................ 277

8 Figures
Figure 1. Current crowding contours for a) LFPAK88 andb) D2PAK packages.......................................................................................... 267
Figure 2. LFPAK internal structure............................................................................... 268
Figure 3. Thermal resistances inside and outside the MOSFET............................... 270
Figure 4. Transient thermal impedance from junction to mounting base as a ...........
function of pulse duration............................................................................ 271
Figure 5. Normalized drain-source on-state resistance factor as a function of ..........
junction temperature..................................................................................... 273
Figure 6. Continuous drain curren as a function of mounting basetemperature.................................................................................................... 275
Figure 7. Source-drain (diode forward) current as a function of source-drain(diode forward) voltage; typical value........................................................ 278
Figure 8. Simplified motor drive and DC-to-DC converter application
diagram............................................................................................................. 279
Figure 9. Output load drive circuit 1 (motor drive)................................................... 279
Figure 10. Output load drive circuit 2 (DC-to-DC)........................................................ 279
Figure 11. Motor operation parameters........................................................................ 280
Figure 12. MOSFETs used as isolators T2 and T3......................................................... 281
Figure 13. Reverse polarity protection MOSFET T1.................................................... 282
Figure 14. Reverse polarity protection using back-to-back MOSFETs...................... 283

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Chapter 11
LFPAK MOSFET thermal
resistance - simulation, test
and optimization of
PCB layout
Application Note: AN90019

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1 Introduction
This chapter explains the parameter thermal resistance from junction to ambientand how it can be reduced by careful PCB layout.
Nexperia MOSFET data sheets contain a parameter Rth(j-a) which is the thermal
resistance from junction to ambient. This is a guide to how much heat can be
dissipated from the device to its surroundings in a typical application. For asurface-mounted device such as an LFPAK MOSFET this is highly dependent on thetype of Printed Circuit Board (PCB) on which it is mounted. The type of material,
the thickness of copper and the shape of the copper footprint all contribute toRth(j-a).
This application note looks at the structure of an LFPAK MOSFET and the way it ismounted on a PCB. It uses thermal modelling techniques to analyse how heat,generated inside the device, is transferred to its surroundings. Various circuit
layouts are considered and tested so that developers can use a value for Rth(j-a) thatclosely reflects practical applications.

2 Definition of thermal resistance Rth(j-a)
Rth(j-a) is the thermal resistance from the active surface of the silicon crystal to thesurrounding environment. It is a value that represents the net effect of all thepossible series and parallel paths from the semiconductor junction to ambient andincludes heat transfer by means such as conduction, convection and radiation.
Rth(j-a) is defined as the temperature difference between junction and ambient thattransfers one watt of power to the environment. It is given by:Rth(j-a) =Tj - Ta__________[1]P
where Rth(j-a) is measured in K/W Tj is the junction temperature (°C)
Tamb is the ambient temperature (°C)
P is the heating power dissipated inside the MOSFET (W)

2.1 Test method
The method used to measure Rth(j-a) is defined by JEDEC standard 51-1 and 51-2A.
The approach is a static implementation, meaning heating power is applied oncontinuous basis while monitoring the junction temperature. The formula for Rth(j-a)
shown above can be used once a known power is applied and steady state is288

reached. The Device Under Test (DUT) is soldered to a FR4 test board and placed in
the geometric centre of the test enclosure as shown in Figure 1 and Figure 2. The
enclosure comprises a 305 mm cube made from low thermal conductivity materials
such as cardboard, polycarbonate, polypropylene, wood, etc.

Dimensions in mm.
Wiring not shown, but will be placed to avoid interferingwith the convection air flow.

Figure 1 | Side view of the test fixture and enclosure

Figure 2 | Isometric view of the test board						

and fixture without the enclosure

2.2 Heat transfer in the JEDEC enclosure
Within the JEDEC enclosure, heat is transferred from the DUT to the PCB andsurroundings by three basic mechanisms; conduction, convection, and radiation.
Conduction is the process by which heat transfers from high temperature regions
to low temperature regions that are in contact with each other. Figure 3 shows howheat is transferred from the semiconductor junction to the copper clip and fromthe clip to the encapsulant. Also, from the junction through the silicon die to themounting base, and from the mounting base to the PCB.
Convection refers to the heat transfer process caused by displacement of air
surrounding the test fixture. The movement is caused by the tendency of hotter
and therefore less dense air to rise, and colder, denser air to sink under the
influence of gravity, which consequently results in transfer of heat.

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Radiation does not depend on contact as heat conduction and convection do.
Thermal radiation is when an object loses energy in the form of electromagnetic
radiation in the infrared part of the spectrum. In the example shown in Figure 3,
heat is radiated from the surfaces of the MOSFET and the PCB into the space inside
the box.

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Figure 3 | LFPAK MOSFET - heat transfer model

The measured thermal resistance of a MOSFET, Rth(j-a), depends upon the package
size, the material properties and the internal structure of the device. It also
depends on the PCB that is used in the test fixture. Modern surface-mount
MOSFETs are designed to use the PCB as a heat sink and rely on the copper traces
to spread the heat over a large area to assist with cooling.
Nexperia power MOSFET data sheets usually specify two values for Rth(j-a) – onewith a minimum footprint that conforms with the JEDEC standard, and anotherwith a 25.4 mm (1") square of 70 µm thick (2 oz) copper on the top surface. See
Table 1 also Figure 4 and Figure 5 below.

Table 1 | Thermal characteristics
Symbol Parameter
Rth(j-mb)

thermal resistance fromjunction to mounting baseRth(j-a)thermal resistance fromjunction to ambient

Conditions

Min

Typ

Max Unit-

0.56 0.63 K/W

Figure 4

-

50

-

K/W

Figure 5

-

125

-

K/W

aaa-005750

aaa-005751

Figure 4 | PCB layout for thermal resistance

Figure 5 | PCB layout for thermal resistancejunction to ambient 1” square pad; 		

junction to ambient minimum

FR4 Board; 2oz copper				footprint; FR4 Board; 2oz copper

3 MOSFET drain-source current pathModern power MOSFETs have an on-state resistance in the order of milliohms andsurface-mount variants are capable of switching hundreds of amps. In these cases
consideration must be given to the Joule heating effect, i.e. induced heat due tocurrent flowing through the conductor, in this case current through the device and
the PCB traces.
Figure 6, below, shows the current path through a LFPAK MOSFET. The clip is
attached directly to the semiconductor junction and at low current it acts as a heatsink, conducting heat away from the junction to the PCB. At high current, however,it becomes heated by its own resistance and requires cooling by being connected
to a cooling surface connected to the source terminal.

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Figure 6 | LFPAK package MOSFET current path291

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4 PCB layout optimization
4.1 PCB trace width calculation
To examine the heating effect of the PCB trace let us consider as an example a
Nexperia LFPAK MOSFET; PSMN1R0-30YLD. This has an on-state resistanceRDS(on) = 1 mΩ (VGS = 10 V; Tj = 25 °C).
At a typical current of 55 A, the simplified power dissipation, given by I2R will be
552 x 0.001 = 3 W. This power can be dissipated with the appropriate PCB layout
applying the right amount of copper to spread the heat.
Many PCB manufacturers websites include a trace width calculation tool based on
the IPC-2221 Generic Standard on Printed Board Design. Using an IPC-2221 trace
width calculator we can determine the size of trace required to carry 55 A.
Assuming 70 µm (2 oz) thick copper and allowing for manageable 30 °C rise in the
trace temperature, Figure 7 shows that a trace almost 20 mm wide is required.

Figure 7 | Example IPC-2221 trace width calculator292

4.2 PCB layout simulations
A number of thermal simulations have been carried out using a PSMN1R0-30YLD as
an example to find the optimum PCB layout that extracts the maximum amount ofheat from the device and therefore gives the lowest possible value for Rth(j-a). All
the simulations were on a 25.4 mm square of 70 µm thick copper. The MOSFET was
fully turned on with a constant current that produced a 3 W power loss in thedevice.
The investigation examined different positions of the MOSFET within the copper
square. It also considered allocating different percentages of the copper square to
the source and drain.
Finally, it examined the heating effect of trace width on the temperature of the
MOSFET and the consequences for Rth(j-a) measurement.

Test layout 1
In Figure 8 the device is mounted on the edge of a 25.4 mm square copper pad.
Rth(j-a) = 41.8 K/W according to the simulation result.
This layout assumes that the main heat transfer path is from the junction, throughthe silicon die to the mounting base. All of the copper pad is allocated to coolingthe mounting base. We see from the thermal profile that very little heat reaches
parts of the copper furthest away from the MOSFET and these contribute little tothe cooling. The source trace is very hot because of inefficient cooling of the source
pins. This is evidently not an optimum solution resulting in a relatively high value
for Rth(j-a).

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Figure 8 | Test layout 1 and surface temperature profile

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Test layout 2
The test layout of Figure 9 shows the MOSFET mounted in the middle of the25.4 mm copper square with the copper area divided equally between source anddrain.
The simulation result gives: Rth(j-a) = 39.1 K/W
Placing the device in the middle of the copper has improved the cooling andreduced the thermal resistance. However, the copper area connected to the drain
tab is hotter than that connected to the source pins which suggest that more heatflows through the drain tab. Selecting the proper ratio of source to drain copper isimportant to obtain the optimum Rth(j-a) value.

Figure 9 | Test PCB layout 2 and surface temperature profile

Test layouts 3, 4 and 5
In Figure 10, Figure 11and Figure 12 different layouts examine the effect of various
trace widths for the source connection.
Layout 3 has the MOSFET located in the centre of the copper pad and the Joule
heating effect of the source trace is not considered.
The simulated result gives: Rth(j-a) = 38.6 K/W.

294

Figure 10 | Test PCB layout 3 and surface temperature profile

However in a practical working circuit when low RDS(on) MOSFETs are fitted, thedrain-source current can reach tens or even hundreds of amperes, and the heatgenerated in the PCB traces cannot be neglected.
Layout 4 shows what happens when the Joule heating of the source trace is taken
into account. In this case the trace width is 0.8 mm. The power loss in the MOSFET
is 3 W as before, but the power loss in the trace is 8.8 W. Figure 11 shows the rise intemperature of the source trace which is higher than the die temperature of the
MOSFET and is therefore heated significantly.

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Figure 11 | Test PCB layout 4 and surface temperature profile

Layout 5 has the source trace increased to 3 mm. The test conditions are the sameas in layout 4. This time the power loss in the source trace is reduced to 2.4 W but
Figure 12 shows that the temperature still exceeds that of the die.

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The simulated result gives: R1111= 56.7 K/W.

Figure 12 | Test PCB layout 5 and surface temperature profile

We can deduce from test layouts 4 and 5 that in a practical application involving an
LFPAK MOSFET careful consideration must be given to trace thickness and width sothat the Joule heating of the current path does not contribute to the heating of
the MOSFET.
Optimal PCB layout - layout 6
Layout 6, shown in Figure 13, attempts to optimise the copper traces to obtain thelowest possible Rth(j-a). Approximately 1/3 of the 25.4 mm copper pad is allocated tothe source pins whilst the width of the source trace is a maximum.
The simulation result gives: Rth(j-a) = 37.1 K/W.

Figure 13 | Test PCB layout 6 and surface temperature profile296

5 Conclusion
Nexperia LFPAK MOSFETs are designed to handle high levels of current, 300 A inthe case of PSMN1R0-30YLD and in excess of 400 A for packages such as LFPAK88.
To take full advantage of the MOSFET's ability to handle these levels of current,
designers are faced with the challenge of dissipating the resulting power throughthe PCB and therefore care must be taken in the design of the layout.
A larger amount of copper and a larger area used under the MOSFET's drain tab will
provide better thermal performance and thus better power dissipation. Perhaps
less obvious is the PCB layout/ trace for source pins and the importance of theamount/area of copper used. LFPAK packages provide a good thermal path fromthe junction to the source pins as they use one piece copper-clip (i.e. not two piecesof copper soldered together or copper-ribbon combination). It is therefore very
important, for an efficient thermal design to consider a layout with the rightamount and area of copper for the source pins.
As shown in this application note the best Rth(j-a) can be achieved when this layout
approach is applied. Another important factor to consider is the width andthickness necessary for PCB traces providing the electrical connections to the
MOSFETs, in order to handle the high currents.

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LFPAK Vs DPAK thermal
performance comparison

Addressing thermal
concerns LFPAK88 Vs
D2PAK

Increasing thermal
capability in small signal
MOSFETs

Visit blog

Visit blog
LFPAK56D taking theheat out of engine
management systems

Weighing the benefitsof LFPAK

Using Nexperia PowerMOSFETs to handle highcurrents up to 380 A

High current 3 phase
BLDC motor drive
application using
Nexperia LFPAK88
MOSFETsLFPAK88 A very coolcustomer6 Tables
Table 1. Thermal characteristics.....................................................................................290

7 Figures
Figure 1. Side view of the test fixture and enclosure...............................................289
Figure 2. Isometric view of the test board and fixture
without the enclosure....................................................................................289
Figure 3. LFPAK MOSFET - heat transfer model.........................................................290
Figure 4. PCB layout for thermal resistance junction toambient 1” square pad; FR4 Board; 2 oz copper.......................................291
Figure 5. PCB layout for thermal resistance junction toambient minimum footprint; FR4 Board; 2 oz copper.............................291
Figure 6. LFPAK package MOSFET current path........................................................291
Figure 7. Example IPC-2221 trace width calculator...................................................292
Figure 8. Test layout 1 and surface temperature profile.........................................293
Figure 9. Test PCB layout 2 and surface temperature profile.................................294
Figure 10. Test PCB layout 3 and surface temperature profile.................................295
Figure 11. Test PCB layout 4 and surface temperature profile.................................295
Figure 12. Test PCB layout 5 and surface temperature profile.................................296
Figure 13. Test PCB layout 6 and surface temperature profile.................................296

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Chapter 12
H-bridge motor controller
design using Nexperia
discrete semiconductors
and logic ICs
Application Note: TN90002

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1 Introduction
An example of a H-bridge motor controller designed with Nexperia discrete and
Nexperia logic IC components.
This chapter demonstrates a H-bridge motor controller PCB, built using Nexperia
discrete semiconductors and logic ICs.
The H-bridge circuit is a full bridge DC-to-DC converter allowing operation of abrushed DC motor (48 V max, 12 V min, 5 A max). The key feature of this design isthat all electronic functions are designed with Nexperia discrete and logic IC
components (low cost, no micro-controller or software needed).
+V

+V
Q1

Q2

M
Q3

Q4

aaa-030956

Figure 1 | Simplified MOSFET H-bridge motor control

The left MOSFETs of the full bridge (Q1 and Q3 in the simplified diagram above) are
the switching MOSFETs (see the PCB top view in Figure 26), the right MOSFETs (Q2and Q4 in the simplified diagram above) select the motor rotation direction:• right high side MOSFET fully ON = Forward• right low side MOSFET fully ON = Reverse A switch selects the motor rotation
direction.
Jumpers select one of 3 switching frequencies: 7.8 kHz, 15.6 kHz or 31.3 kHz.
Two tactile push buttons allow the duty cycle (motor speed) to be increased or
decreased. There are 8 steps from 0 to 100% duty cycle. A current limitation
protection avoids over current in the motor and MOSFETs (set at approximately
6.5 A).
This H-bridge motor controller PCB allows the user to choose between 3 Nexperia

302

MOSFET packages (LFPAK33, LFPAK56D or LFPAK56), jumpers are used to connect
the MOSFETs chosen by the user.
This chapter describes each of the main functions used in the design.

2 Block diagram and system functionality
Clock

Power
SupplyPWM

Driving
Circuitry

Overcurrent
Detection

H-bridge

Motoraaa-030969

Figure 2 | H-bridge motor controller block diagram

2.1 Subsystems overview
1. Power Supply• Accepts 12 V to 48 V DC input• Transient overvoltage protection• Reverse polarity protection• Buck converter (12 V)• Linear regulator (5 V) for logic devices
2. Clock and duty cycle generator• 4 MHz crystal oscillator and frequency divider to create 3 different 		
switching frequencies• Duty cycle sets by push button inputs to select the duty cycle (0% to 100%)• 62.5 kHz output is used to supply the charge pump on the driving
		 circuitry block
3. PWM• Reset function to activate other function when VCC 5 V supply is stable• Dead-time function and PWM enable (over current protection disabling)• Level shifter• Direction selection
4. Driver circuit• High-side and low-side drivers to drive 4 MOSFETs of the full bridge• Charge pump to supply the high-side MOSFETs

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5. H-bridge• Reservoir and decoupling capacitors• Snubber on the left MOSFETs (switching MOSFETs)• Jumper to connect the selected MOSFETs• Gate drive resistors• Low-side current measurement
6. Overcurrent detection• Comparator, with voltage reference setting current limit• PWM reset function reactivating PWM if fault disappears

3 Subsystem descriptions
3.1 Power supply
The H-bridge motor controller power supply circuit comprises of:• DC input stage with transient overvoltage and reverse voltage protection• 12 V output DC-to-DC buck converter stage• 5 V output linear regulator (for logic ICs) with status LED

, ,

, ,

aaa-030910

Figure 3 | Power supply circuit

Transient overvoltage protection
D8 and D17 (PTVS60VS1UTR) are transient voltage suppressor diodes, rated at 60
V, 400 W for a 10/1000 μs current pulse waveform. They protect against positive
and negative transients.
Note: These TVS diodes provide protection for transient overvoltage only - notfor DC overvoltage.

304

aaa-030914

Figure 4 | Transient overvoltage protection

Reverse polarity protection
Two parallel P-channel MOSFETs Q1 and Q2 (BUK6Y33-60P) form the reverse
polarity protection.
Q1 and Q2 are biased on through D1 and D2 (BZX84J-B10) respectively when the
supply polarity is positive (VGS= -10 V). Q1 and Q2 are off when the supply is
negative (VGS = VF = 0.7 V), current cannot flow because the MOSFET body diodes
are reverse biased.
This means that if the supply is inverted, no current will flow into the circuit andpotentially cause damage.

, ,

, ,

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aaa-030911

Figure 5 | Reverse polarity protection305

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Buck converter (12 V)
The next stage is a switching regulator that outputs 12 V. To start the switchingregulator a start- up circuit is used consisting of: Zener diode D18 (BZX84J-B5V6),
R10, and C14. This provides a 5 V supply for inverting Schmitt trigger U21(74HC1G14) before VCC (5 V) is available through D19 (BAS316). U21 can then start
the switching regulation.
The buck converter consists of: Q5 (BUK6Y33-60P), Schottky diode D6(PMEG10030ELP) and inductor L1. The output of the buck converter supplies the
12 V rail.
When the output of U21 is high it turns on Q5 via the NPN transistor Q3 (BC846)
and the NPN/ PNP dual transistor pair Q4 (BC846BPN). The voltage on node “V_12”
will then increase. When “V_12” reaches 12 V, D3 ( BZX84J-B10) is conducting andthere is enough voltage on U21 input to force U21 output low. Then Q5 is switched
off and the voltage at “V_12” decreases until U21 input is low enough to restart a
new cycle.

aaa-030912

Figure 6 | Buck converter

Linear regulator (5 V) for logic ICs
A linear regulator comprising NPN transistor Q6 (BCP55), Zener diode D7(BZX84J-B5V6) and R11 provides a regulated VCC (5 V) rail for the logic ICs.

aaa-030913

Figure 7 | Linear regulator

3.2 Clock and duty cycle generator
The Clock circuit comprises of:• Crystal oscillator and frequency divider• Duty cycle generator• Frequency selection jumpers
Frequency selection jumpers

Clock

Duty cycle generator

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aaa-030912

Figure 8 | Clock, duty cycle and frequency selection307H-bridge motor controller design using Nexperia discrete semiconductors and logic ICs

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Crystal oscillator and frequency divider
A crystal oscillator is used for accuracy and stability. The oscillator frequency is 4MHz.
U1 (74HCT4060) is a frequency divider, used to obtain different frequencies fromthe 4 MHz. It provides outputs at:• 62.5 kHz to drive the charge pump circuit (see Figure 19)• 31.3 kHz, 15.6 kHz and 7.8 kHz for the switching frequency (user selectable)
The switching frequency is selected by the multiplexer U3 (74HCT151) according to
the jumper settings of JP7, JP8 and JP9.
Note: Only one jumper should be fitted at the same time.
The switching frequency output is labeled “Clock”.
A reset signal enables both multiplexer U3 and U19 (see next section for U19
explanation) when VCC is 5 V.

aaa-030937

Figure 9 | Oscillator and frequency divider

Duty cycle generator
Signals with a frequency 8 times higher than the required final switching frequencies are fed to 3 of the inputs of an 8-bit multiplexer IC, U19 (74HCT151). These arelabeled “8*Frequency_duty_cycle step”.
Jumpers JP7, JP8, JP9 select both the switching frequency “Clock” and at the sametime, “Frequency_duty_cycle step” which is a multiple (x8) of the switchingfrequency.

308

This higher frequency is used to create 8 duty cycle steps.
“Frequency_duty_cycle step” goes into the clock input of programmable timer U18(74HC40103PW). U18 uses "Frequency_duty_cycle step” as a clock.
U18 output produce a duty cycle time in “duty_cycle_output” equals to the value
read in input (P0 to P3) multiplied by the clock period.
"duty_cycle_output” is the general output of the Clock circuit.

aaa-030938

Figure 10 | Duty cycle

Table 1 | Duty cycle selection
P0 (1)

P1 (2)

P2 (4)

P3 (8)

Duty Cycle

0

0

0

0

0%

1

0

0

0

12.5%

0

1

0

0

25%

1

1

0

0

37.5%

-

-

-

-

-

1

1

1

0

87.5%

0

0

0

1

100%

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The signal “Clock” from U3 resets U18 at each new switching period. When reset,
U18 will read inputs P0 to P3 and output a duty cycle in “duty_cycle_output”
proportional to the settings.
The outputs of a 4-bit synchronous binary up/down counter U11 (74HC193) set the
duty cycle value, it is the input of U18. Each time a rising edge appears on the UP309

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input, the output value (Q0 to Q3) increases by 1, each time a rising edge appearson the DOWN input, output value (Q0 to Q3) decreases by 1.
The push buttons SW3 and SW1 with U10 and U22 (74HC1G125) create the risingedges on UP or DOWN input of U11, when the user releases the push button.
When U11 output reaches a maximum count of 9 (8 to 9 transition), (Q0 and Q3=1), the “stop_up” output of U12 (74HC1G08) is set to 1 and U10 (74HC1G125) isdeactivated (high impedance), any further impulses on SW3 can’t produce a risingedge on UP input (U11) and the duty cycle reaches a maximum.
When U11 output reaches a minimum of 0 (Q0 to Q3 = 0), U17 (74HC1G08)“stop_down” output is set to 1 and U22 (74HC1G125) is deactivated (highimpedance), any further impulses on SW1 can’t produce a rising edge on DOWN
input (U11) and the duty cycle reaches a minimum (after 0 the next counter value is15, we need to stop the counter from setting duty cycle to a maximum).

3.3 Pulse Width Modulation
The PWM circuit consists of:••••

PWM enable and reset
Dead-time control
Level shifting stage
Direction selection

aaa-030939

Figure 11 | PWM, dead-time, level shifting and direction selection

310

Reset function
The reset function to activate other function when the VCC (5 V) supply is stable.

aaa-030940

Figure 12 | Reset circuit

When VCC (5 V) reaches approximately 4 V, “Reset_signal” will go high after a delay
set by C15. This validates all functions enabled by this signal. This ensures that all
logic ICs will start with a clean VCC (5 V).
Adjustable precision shunt regulator D22 (TL431BCDBZR) acts like a comparator.
PNP transistor Q36 (BC857A) is a current source charging C15 when the thresholdis reached.
MOSFET Q37 (2N7002) discharge C15 when VCC (5 V) is no longer present.
The dual NPN/PNP resistor-equipped transistor Q38 (PUMD13) switches “Reset_
signal” high when the voltage on C15 (delay setting) reaches a value high enoughto turn on the NPN transistor (input, pin 2).
Dead-time function and PWM enable (over current protection disabling)“duty_cycle_output” is the high side command (α) and “duty_cycle_output" isinverted by U20 (74HC1G14) to create the low side command (1-α).

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The duty cycle command is disabled by U6 (74HC125) through its input “PWM_EN”
equal to 1 (over current protection switch U6 output in high impedance).
A dead time is controlled by U2 (74HC14) and RC filter. Q39, Q40 (2N7002) force
the duty cycle to 0 when “PWM_EN” is 1.

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aaa-030941

Figure 13 | Dead-time function and PWM enable

Level shifter
Transistors Q9 and Q10 (BC846) interface the 5 V duty cycle signal to the high side
and low side drivers (see Section 3.4)

aaa-030942

Figure 14 | Level shifter312

Direction selection
Switch SW2 selects the motor direction by activating either Q7 or Q8 (BC846BS).

aaa-030943

Figure 15 | Direction selection

Q7 or Q8 are fully ON and activate one of the right MOSFETs: high side fully ON or
low side fully ON.

3.4 Drive circuit
The drive circuit comprises of high-side and low-side drivers to drive the 4 power
MOSFETs that form the full bridge.

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MOSFET & GaN FET Application Handbook

aaa-030944

Figure 16 | Driver circuit313

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Nexperia | A Power Design Engineer's Guide

High-side and low-side driver
Two high side drivers supplied by the charge pump function and regulated at 9 V:
• Q19 (BC846), D15 (BZX84J-B10), C23, R45, Q21(BC846BPN) driven by “HS_Drive_R” for the right high side
• Q20 (BC846), D16(BZX84J-B10), C24, R46, Q22(BC846BPN) driven by “HS_Drive_L” for the left high sideaaa-030945

Figure 17 | High side driver circuit

Two Low side drivers supplied by the 12 V from the buck converter and regulated
at 9 V• Q16 (BC846), D11(BZX84J-B10), C18, R40 for the supply:• Q18(BC846BPN) driven by “LS_Drive_L” for the left low side
• Q23 (BC846BPN) driven by “LS_Drive_R” for the right low sideaaa-030946

Figure 18 | Low side driver circuit

314

Charge pump to supply high side MOSFETs
A simple bootstrap function could be used to supply the high side MOSFETs but
due to the fact the duty cycle could be 100% there is no time in this configuration
to recharge the bootstrap capacitor if the high side MOSFET is fully ON (100% duty
cycle) and the low side MOSFET is fully OFF.
The solution is to use a charge pump to supply the high side MOSFETs in every duty
cycle configuration.

aaa-030947

Figure 19 | Charge pump

“C_pump_refresh” from the clock function (see section Section 3.2) controls Q15(BC846),Q17 (BC846BPN) with a frequency of 62.5 kHz, 50% duty cycle. The 9 V regulated
from “LS_Supply_Rail” is applied on C19 bottom pin at 62.5 kHz, during 50% of thetime, and 0 V during other 50% of the time.
Due to the fact, there is no fast voltage variation across a capacitor, there is always
a voltage equals to “V_SUPPLY” across C19 (“V_SUPPLY” is the supply voltage of the
H-bridge).

H-bridge motor controller design using Nexperia discrete semiconductors and logic ICs

12

MOSFET & GaN FET Application Handbook• When the bottom pin of C19 is 0 V, voltage across C19 is “V_SUPPLY”.
• When the bottom pin of C19 is 9 V, voltage across C19 is “V_SUPPLY” +9 V. C21
and C22 are the filtering capacitor for the driver input voltage.

315

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3.5 H-bridge
The H-bridge circuit comprises of:••••Power MOSFETs
Reservoir and decoupling capacitors
Snubber components
Low side current measurement

, ,

, ,

, ,

, ,

,

,

7, 8

7, 8

, ,

, ,

, ,

, ,

aaa-030948

Figure 20 | H-bridge

Reservoir and decoupling capacitors
Decoupling capacitor (1 μF) and reservoir capacitor (220 μF) are used to filter the“V_SUPPLY”.
Snubber on the left MOSFETs (switching MOSFETs)
RC Snubber footprint are implemented on the switching MOSFETs.
An additional RC snubber may be added to further reduce ringing, application note
AN11160 (chapter 6 in this book) - Designing RC snubbers gives an explanation
about snubber calculation.

, ,aaa-030949

Figure 21 | Snubber circuit

Jumper to connect the required MOSFETs
Each MOSFET package is selected by soldering the following jumpers:• LFPAK33: JP3, JP4• LFPAK56D: JP1, JP6• LFPAK56: JP2, JP5
Note: Solder only the jumpers needed to avoid incorrect operation!

Gate drive resistors
To avoid conflict between the switching MOSFETs gate resistors should be
desoldered for those MOSFETs not in use.
Adjust the value of the gate resistor to reduce ringing or modify the switching time.
A resistor between gate and source of each MOSFET keeps the MOSFET off when
there is no gate voltage applied.

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Low side current measurement

Current
measureaaa-030950

Figure 22 | Current measurement

A shunt resistor R16 (20 mΩ) measures the current in the H-bridge and provide a
voltage proportional to the current to the over current protection input ”IN-“. A low
pass filter R17, C13 filters the signal to avoid false triggering of the current limit
circuit due to the noise spike.

3.6 Overcurrent detection
Overcurrent in the motor drive MOSFETs is prevented by comparing the current
sense voltage with a reference voltage. In the case of overcurrent the PWM
function is reset. The PWM function will be reactivated if the fault disappears.

PWM reset

Comparator - current protection

aaa-030951

Figure 23 | Overcurrent detection

Comparator, with voltage reference setting current limit
Comparator - current protection

aaa-030952

Figure 24 | Comparator, with voltage reference

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12

MOSFET & GaN FET Application Handbook"IN+” is the maximum current setting input. “IN-“ is the current measurement input
reference. R35 is a rudimentary current source for the comparator circuit.
Q13 (BCV61) is a current mirror, this means that the polarisation current from R35
is shared equally between the two current paths.
Q11 and Q12 (PMP5501Y) are configured as Darlington pairs to increase the gain
to improve the sensitivity of the comparator.

319

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Nexperia | A Power Design Engineer's Guide

Q11 is a differential pair.
• When IN+ > IN- the output (TP28) is high (no overcurrent)• When IN+ < IN- the output (TP28) is low (overcurrent)
R33 and R34 set the current limitation, R32 adds hysteresis to avoid transitions dueto the noise.
PWM reset
The PWM reset function reactivating the PWM circuit if fault disappears.
PWM reset

aaa-030953

Figure 25 | PWM reset

When there is no overcurrent TP28 is high. “Reset_signal” is high since the VCC (5 V)
is supplying the circuit. The output of AND gate U4 (74HC1G08) is high.
The signal “Clock” makes “PWM_EN” output from U8 (74HC74) switch to 0 at thefirst rising edge and stay at 0. U2 (74HC14) create a delay to have D input highbefore the rising edge on Cp).
If an overcurrent is detected by the comparator TP28 will switch to 0, TP29 will also
switch to 0. Instantaneously “PWM_EN” will switch to 1 and force the duty cycle tobe 0 (see section Dead-time function).
When the overcurrent disappears, TP28 and TP29 switch back to 1 and the next
rising edge of “Clock” makes “PWM_EN” switch to 0 and the duty cycle is reenabled.
This allows a cycle by cycle current limit behaviour from the circuit.

320

4 PCB top viewnote: gate resistors R57,R58, R78, R79 and jumpers JP2, JP5 are fitted, selecting the MOSFETs in LFPAK56 package

Figure 26 | H-bridge motor controller PCB top-view

5 Schematics
Power Supply

H-bridge
Driver circuitry

Clock
PWM

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MOSFET & GaN FET Application Handbook

Comparator

Decoupling capacitors

aaa-030955

Figure 27 | Schematics overview

321

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5.1 Power supply
DC Supply 12 V - 48 V

Linear regulator
12 V to 5 V

Buck converter 12 V output

, ,

, ,

LED 5 V OK

Reverse polarity protection

aaa-030910

Figure 28 | Power supply schematic

5.2 Clock
Frequency selection jumpers

Clock

aaa-030937

Figure 29 | Clock schematic

5.3 Duty cycle
Duty cycle generator

aaa-030938

Figure 30 | Duty cycle schematic

5.4 PWM
PWM enable and reset

Dead-time control

Level shifterH-bridge motor controller design using Nexperia discrete semiconductors and logic ICs

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MOSFET & GaN FET Application Handbook

aaa-030939

Figure 31 | PWM schematic323

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Nexperia | A Power Design Engineer's Guide

5.5 Overcurrent detection
Unused components

PWM reset

Comparator - current protection

aaa-030954

Figure 32 | Overcurrent detection schematic

5.6 H-Bridge
H-bridge

, ,

, ,

, ,

, ,

,

,

7, 8

7, 8

, ,

, ,

, ,

, ,

Current
measureaaa-030948

Figure 33 | H-brdige schematic324

5.7 Driver circuit
H-bridge driving circuit

Low-side driving circuit

aaa-030944

Figure 34 | Driver circuit schematic

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H-bridge motor controller design using Nexperia discrete semiconductors and logic ICs

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Nexperia | A Power Design Engineer's Guide

6 Bill of materials (Nexperia parts)
Table 2: BOM (Nexperia parts)
Reference

Part number

Description

D1, D2, D3, D5, D11, D15,
D16

BZX84J-B10Single Zener diode 10 V

D4, D9, D10, D19

BAS316High-speed switching diodes

D6

PMEG10030ELP

100 V, 3 A low leakage
current Schottky barrier
rectifier

D7, D18, D20, D21

BZX84J-B5V6

Single Zener diodes 5.6 V

D8, D17

PTVS60VS1UTR

High-temperature 400 W
Transient Voltage Suppressor

D12, D13, D14

PMEG6010ER

1 A low VF MEGA Schottky
barrier rectifier

D22

TL431BCDBZR

Adjustable precision shunt
regulator

Q1, Q2, Q5

BUK6Y33-60P

P-channel MOSFET in an
LFPAK56 (Power SO8)
package 60 V, 33 mΩ

Q3, Q9, Q10, Q15, Q16, Q19,
BC846
Q20

65 V, 100 mA NPN generalpurpose transistors

Q4, Q17, Q18, Q21, Q22,
Q23

BC846BPN

65 V, 100 mA NPN/PNP
general-purpose transistor

Q6

BCP55

60 V, 1 A NPN medium powertransistor

Q7, Q8BC846BS65 V, 100 mA NPN/NPN
general-purpose transistor

Q11, Q12

PMP5501Y

PNP/PNP matched doubletransistors

Q13

BCV61

NPN general-purpose doubletransistors

Q14

BC847BS45 V, 100 mA NPN/NPN
general-purpose transistor

Q24, Q33

BUK7K12-60E

LFPAK56D, dual N-channel
MOSFETReferencePart number

Description

Q25, Q26, Q31, Q32

BUK7Y7R8-80E

LFPAK56, N channel MOSFET,
80 V, 7.8 mΩ

Q27, Q28, Q29, Q30

BUK7M9R9-60E

LFPAK33, N channel MOSFET

Q34, Q35, Q37, Q39, Q40

2N7002

60 V, 300 mA N-channel
Trench MOSFET

Q36

BC857APNP general purposetransistors

Q38

PUMD13

NPN/PNP resistor-equipped
transistors; R1 = 4.7 kΩ, R2 =
47 kΩ

U1

74HC4060PW

14-stage binary ripple
counter with oscillator

U2

74HC14PW-Q100

Hex inverting Schmitt trigger

U3, U19

74HC151PW-Q100

8-input multiplexer

U4, U12, U17

74HC1G08GW-Q100

2-input AND gate

U6

74HC125PW-Q100

Quad buffer/line driver;3-stateU8

74HC74PW-Q100

Dual D-type flip-flop with set
and reset; positive-edge
trigger

U9, U20, U21

74HC1G14GW-Q100

Inverting Schmitt trigger

U10, U22

74HC1G125GW-Q100

Bus buffer/line driver; 3-state

U11

74HC193PW-Q100

Presettable synchronous
4-bit binary up, down counter

U13, U14

74HC1G02GW-Q100

2-input NOR gate

U15, U16

74HC1G00GW-Q100

2-input NAND gate

U18

74HC40103PW

8-bit synchronous binary
down counter

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7 Bill of materials
Table 3: BOM
Item Qty

Reference

Value

Footprint

1

20

C1, C2, C3, C4, C5,
C6, C7, C8, C9, C18,
C20, C23, C24, C39,
C40, C44, C45, C46,
C48, C49

1 µF

Capacitors_SMD:
C_1206

2

1

C10

47 µF

Capacitors_SMD: CP_
Elec 6.3 x 5.7

3

3

C11, C12, C43

51 pF

Capacitors_SMD:
C_08054

1

C13

100 pF

Capacitors_SMD:
C_0805

5

3

C14, C41, C42

1 µF

Capacitors_SMD:
C_08056

1

C15

470 nF

Capacitors_SMD:
C_0805

7

2

C16, C17

2.2 nF

Capacitors_SMD:
C_0805

8

10

C19, C21, C22, C26,
C27, C34, C35, C36,
C37, C38

1 µF

Capacitors_SMD:
C_1206

9

2

C25, C47

220 µF"Capacitors_THT:
100 V
CP_Radial_D 12.5 mm P

106C28, C29, C30, C31,
C32, C33* user selected

5.00 mm"11

10

C50, C51, C52, C53,
C54, C55, C56, C57,
C58, C59

1 µF

Capacitors_SMD:
C_080512

7

D1, D2, D3, D5, D11,
D15, D16

BZX84J-B10

Capacitors_SMD:
C_080513

4

D4, D9, D10, D19

BAS316141

D6PMEG10030ELP

Diodes_SMD:
D_SOD-323
HandSoldering
Diodes_SMD:
D_SOD-323
HandSoldering

Voltage

100 V

100 V

Item Qty

Reference

Value

Footprint

15

4

D7, D18, D20, D21

BZX84J-B5V6

Nexperia: SOD128

16

2

D8, D17

PTVS60VS1UTR

Diodes_SMD:
D_SOD-323
HandSoldering

17

3

D12, D13, D14

PMEG6010ER

Nexperia:
D_SOD123W

18

1

D22

TL431BCDBZR

Nexperia:
D_SOD123W

19

1

D23

LED

TO_SOT_Packages_
SMD: SOT-23

20

1

J1

Conn_01x02

LEDs: LED_0603
Connectors Phoenix:
PhoenixContact_
GMSTBVA-G_ 02 x 7.50mm Vertical

21

1

J4

Conn_01x02_
FemaleConnectors Phoenix:
PhoenixContact_
GMSTBVA-G_ 02 x 7.50mm Vertical

22

9JP1, JP2, JP3, JP4,
JP5, JP6, JP7, JP8,
JP9

JumperJumpers: SolderJumper2_P 1.3 mm Open_
TrianglePad 1.0 x 1.5
mm

23

1

L1

390 µH

Inductors_THT: L_
Radial_D 10.0 mm P
5.00 mm Fastron_07M

24

4

MK1, MK2, MK3, MK4 Mounting_Hole

Mounting_
Holes:MountingHole 4.3
mm M4

25

3

Q1, Q2, Q5

BUK6Y33-60P

TO_SOT_Packages_
SMD: SOT-669_LFPAK

26

7

Q3, Q9, Q10, Q15,
Q16, Q19, Q20

BC846

TO_SOT_Packages_
SMD: SOT-23

27

6

Q4, Q17, Q18, Q21,
Q22, Q23

BC846BPN"TO_SOT_Packages_
SMD: SOT-363_SC-70-6
Handsoldering"

28

1

Q6

BCP55

TO_SOT_Packages_
SMD: SOT-223 3Lead
TabPin2

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Item Qty

Reference

Value

Footprint

29

2

Q7, Q8BC846BS"TO_SOT_Packages_
SMD:
SOT-363_SC-70-6
Handsoldering"30

2

Q11, Q12

PMP5501Y"TO_SOT_Packages_
SMD:
SOT-363_SC-70-6
Handsoldering"

31

1

Q13

BCV61

Nexperia: SOT-143B

32

1

Q14

BC847BS"TO_SOT_Packages_
SMD: SOT-363_SC-70-6
Handsoldering"33

2

Q24, Q33

BUK7K12-60E

Nexperia: LFPAK56D

34

4

Q25, Q26, Q31, Q32

BUK7Y7R8-80E

TO_SOT_Packages_
SMD: SOT-669_LFPAK

35

4

Q27, Q28, Q29, Q30

BUK7M9R9-60E

Nexperia: LFPAK33

36

5

Q34, Q35, Q37, Q39,
2N7002
Q40

TO_SOT_Packages_
SMD: SOT-23

37

1

Q36

BC857A

TO_SOT_Packages_
SMD: SOT-23

38

1

Q38

PUMD13

TO_SOT_Packages_
SMD: SOT-363 SC-70-6
Handsoldering

39

2

R1, R2

10 kΩ

Resistors_SMD: R_1206

40

14

R3, R11, R17, R19,
R23, R29, R30, R34,
R39, R40, R42, R48,
R50, R87

1 kΩ

Resistors_SMD: R_0805

34

R4, R6, R12, R13, R14,
R20, R24, R36, R37,
R44, R45, R46, R47,
R49, R51, R52, R55,
R56, R59, R62, R63,
10 kΩR70, R71, R76, R77,
R81, R82, R83, R86,
R88, R89, R91, R92,
R104

Resistors_SMD: R_0805

41

330

VoltageItem Qty

Reference

Value

Footprint

42

10

R5, R7, R18, R21, R22,
R27, R28, R31, R38,
4.7 kΩ
R41

Resistors_SMD:R_0805

43

6

R8, R60, R66, R68,
R69, R74* user selected

Resistors_SMD:R_0805

44

3

R9, R108, R109

100 Ω

Resistors_SMD: R_0805

45

1

R10

20 kΩ

Resistors_SMD: R_1206

46

4

R15, R43, R80, R93

1 MΩ

Resistors_SMD: R_0805

47

1

R16

20 mΩ

Resistors_SMD: R_2512

48

3

R25, R26, R101

2.2 kΩ

Resistors_SMD: R_0805

49

4

R32, R61, R67, R96

100 kΩ

Resistors_SMD: R_0805

50

1

R33

51 kΩ

Resistors_SMD: R_0805

51

1

R35

47 kΩ

Resistors_SMD: R_0805

52

17

R53, R54, R57, R58,
R64, R65, R72, R73,
R78, R79, R84, R85,
R90, R103, R105,
R106, R107

10 Ω

Resistors_SMD: R_0805

53

1

R75

510 kΩ

Resistors_SMD: R_0805

54

4

R94, R97, R99, R100

1.5 kΩ

Resistors_SMD: R_0805

55

1

R95

2.4 kΩ

Resistors_SMD: R_0805

56

2

R98, R102

5 kΩ

Resistors_SMD: R_0805

57

1

R110

500 Ω

Resistors_SMD: R_0805

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Item Qty

Reference

Value

Footprint

58

2

SW1, SW3

SW_Push"Buttons_Switches_
SMD: SW_DIP_x1_W
8.61 mm_Slide_
LowProfile"

59

1

SW2

SW_DPDT_x2

Buttons_Switches_THT:
SW_CuK_
JS202011CQN_DPDT_
Straight

60

1

U1

74HC4060PW"Housings_SSOP:TSSOP-16 4.4 x 5 mm
Pitch 0.65 mm"

61

1

U2

74HC14PW-Q100

Housings_SSOP:TSSOP-14 4.4 x 5 mm_
Pitch 0.65mm

622

U3, U19

74HC151PW-Q100"Housings_SSOP:TSSOP-16 4.4 x 5 mm
Pitch 0.65 mm"

63

3

U4, U12, U17

74HC1G08GW-Q100

TO_SOT_Packages_
SMD: SOT-353_SC-70-5

64

1

U6

74HC125PW-Q100

Housings_SSOP:TSSOP-14 4.4 x 5mm
Pitch 0.65 mm

65

1

U8

74HC74PW-Q100"Housings_SSOP:TSSOP-14 4.4 x 5 mm
Pitch 0.65 mm"

66

3

U9, U20, U21

74HC1G14GW-Q100

TO_SOT_Packages_
SMD: SOT-353_SC-70-5

67

2

U10, U22

74HC1G125GW- Q100

TO_SOT_Packages_
SMD: SOT-353_SC-70-5

68

1

U11

74HC193PW-Q100

Housings_SSOP:TSSOP-16 4.4 x 5mm
Pitch 0.65 mm

692

U13, U14

74HC1G14GW-Q100

TO_SOT_Packages_
SMD: SOT-353_SC-70-5

70

2

U15, U16

74HC1G14GW-Q100

TO_SOT_Packages_
SMD: SOT-353_SC-70-5

VoltageItem Qty

Reference

Value

Footprint

71

1

U18

74HC40103PW"Housings_SSOP:TSSOP-16 4.4 x 5 mm
Pitch 0.65 mm"

72

1

Y1

CrystalCrystals: Crystal_SMD_
HC49-SD

Visit blog
The right package for 12
48 V DC-to-DC conversion

H-bridge DC motorreference design using
Nexperia components

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8 TablesTable 1.
Table 2.
Table 3.

Duty cycle selection........................................................................................ 309
BOM (Nexperia parts)............................................................................ 326-327
BOM.......................................................................................................... 328-333

9 Figures
Figure 1. Simplified MOSFET H-bridge motor control..............................................302
Figure 2. H-bridge motor controller block diagram..................................................303
Figure 3. Power supply circuit.......................................................................................304
Figure 4. Transient overvoltage protection................................................................305
Figure 5. Reverse polarity protection..........................................................................305
Figure 6. Buck converter................................................................................................306
Figure 7. Linear regulator..............................................................................................307
Figure 8. Clock, duty cycle and frequency selection.................................................307
Figure 9. Oscillator and frequency divider..................................................................308
Figure 10. Duty cycle.........................................................................................................309
Figure 11. PWM, dead-time, level shifting and direction selection..........................310
Figure 12. Reset circuit.....................................................................................................311
Figure 13. Dead-time function and PWM enable.........................................................312
Figure 14. Level shifter.....................................................................................................312
Figure 15. Direction selection.........................................................................................313
Figure 16. Driver circuit....................................................................................................313
Figure 17. High side driver circuit...................................................................................314
Figure 18. Low side driver circuit....................................................................................314
Figure 19. Charge pump...................................................................................................315
Figure 20. H-bridge............................................................................................................316
Figure 21. Snubber circuit................................................................................................317
Figure 22. Current measurement...................................................................................318
Figure 23. Overcurrent detection...................................................................................319
Figure 24. Comparator, with voltage reference...........................................................319
Figure 25. PWM reset.......................................................................................................320
Figure 26. H-bridge motor controller PCB top-view....................................................321
Figure 27. Schematics overview......................................................................................321
Figure 28. Power supply schematic................................................................................322
Figure 29. Clock schematic...............................................................................................322
Figure 30. Duty cycle schematic......................................................................................323
Figure 31. PWM schematic...............................................................................................323
Figure 32. Overcurrent detection schematic................................................................324
Figure 33. H-bridge schematic.........................................................................................324
Figure 34. Driver stage schematic..................................................................................325

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Power and small-signal MOSFET frequently asked questions and answers

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Chapter 13
Power and small-signal
MOSFET frequently
asked questions
and answers
Technical Note: TN00008

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1 Introduction
This chapter provides several important questions regarding the use of MOSFETs
and the platforms required. Although it is focused on automotive applications, theprinciples can apply to industrial and consumer applications. It strives to provide
clear answers to these questions and the reasoning behind the answers.
This chapter is intended for guidance only. Any specific questions from customers
should be discussed with Nexperia power MOSFET application engineers.

2 Gate2.1 Q: Why is the VGS rating of Trench 6
automotive logic level MOSFETs limited to
10 V and can it be increased beyond 10 V?
A: The VGS rating of 10 V given to Trench 6 logic level MOSFETs is driven by our<1 ppm failure rate targets and was rated to the best industry practices at the time.
The ppm failure figures are not given in any data sheet nor are they part of
AEC-Q101 qualification. In other words, two devices can both be qualified to
AEC-Q101 and still have different ppm failure rate figures.
Methods of defining, characterising and protecting these ratings have improved
and there is now a possibility to operate beyond the given rating of 10 V. This will
be a function of time, voltage and temperature. For further explanation see below;
more details are available in Nexperia application note AN90001 (chapter 2 in thisbook) - Designing in MOSFETs for safe and reliable gate-drive operation.
Additional information
There are two key words in the above question that are worth expanding on “rating” and “logic level”.
Logic level MOSFETs are primarily intended for applications where the drive
voltage is 5 V and thus optimised accordingly. To achieve a fully-on MOSFET andbest RDS(on) performance with relatively low gate voltages these MOSFETs willneed a thinner gate oxide than standard-level parts which operate with a driveof 10 V VGS. Thinner gate oxides will breakdown at lower voltages and will havea lower rating than standard level devices, (full details are given in AN90001(chapter 2 in this book) - Designing in MOSFETs for safe and reliable gate-drive
operation.

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However, there are instances where logic level MOSFETs are chosen for
non-logic level applications. For example, in automotive applications where thebattery supply voltage can drop to a level where drive circuits need to operate
below 6 V. Therefore, MOSFETs must turn ON with lower gate voltages thanstandard level MOSFETs are capable of. Conversely, the MOSFET gate needs towithstand the nominal battery voltage of approximately 12 V.
Is a logic level MOSFET suitable?
In terms of capability logic level MOSFETs are not expected to suddenly fail assoon as higher voltages are applied. However, applying higher VGS than the
maximum rated voltage will compromise the <1 ppm failure rate and therefore
Nexperia would not consider including these ratings in data sheets.
Nexperia's methodology of removing defectives and reducing early life failuresis achieved through effective screening at production. As a supplier Nexperia iscommitted to zero defects and high quality levels. Therefore, the ratings may
appear lower than our competitors where commitment to quality is possibly
not as stringent. Nexperia VGS max ratings are based on applying 100% max(rated) voltage at 175 °C for 1000 hours with failure rate <1 ppm – for moredetails see: AN90001 (chapter 2 in this book) - Designing in MOSFETs for safeand reliable gate-drive operation.
Designers must consider failure rates figures for logic level MOSFETs when VGS
in their data sheets is rated to ±20 V
A model exists at Nexperia for calculating life failure rates with higher gate
voltages against temperatures. This information can be provided upon request
as a calculated figure and will only be given as a guide.

2.2 Q: Why is VGS maximum limit of Trench 9
automotive logic level MOSFETs different toprevious generations?

13Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

A: The rating is based on meeting AEC-Q101 requirements, as it is with previous
generations. However, Nexperia has developed a new testing methodology which
ensures <1 ppm failure rates over lifetime at the rated VGS. This has been applied toTrench 9 and its VGS rating is set to meet this new requirement.
Additional information
Detailed explanation can be found in AN90001 (chapter 2 in this book) - Designingin MOSFETs for safe and reliable gate-drive operation..

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3 Thermals
3.1 Q: When comparing Zth curves in some data
sheets, there appear to be some
contradictions. From an Rth point of view, theBUK9Y38-100E (Trench generation 6) looks
better (lower). However, from a Zth (at lessthan 100 ms) point of view, the BUK9Y30-75B(Trench generation 3) looks better. The shapeof the graphs indicates that a more advanced
model or measurement was done on the Trench
generation 6 part. Is this assumption correct?
A: The method for setting the Zth curve has changed between the Trench
generation 3 parts (2008) and Trench generation 6 (2012). The die size is also
different which changes the Zth and Rth characteristics.
Additional information
The earlier method used empirical models for Zth (1 μs) and Rth, joined by an
exponential line.
The latest method uses models for the Zth generated by Computational Fluid
Dynamics (CFD) simulation, verified by measurement.
The dies in the 2 parts have different dimensions and therefore they havedifferent Zth.
The plots depicted in Figure 1 compare the data sheet curves for the single
shot Zth. There is a good match between the limit lines for both parts. The
biggest difference is in the region 1 ms to 20 ms.
The conclusion from this comparison, is that the Trench generation 3 part isdesigned to work within these Zth limits. The Trench generation 6 part is an
excellent alternative that has a very high probability of working satisfactorily.
It is possible to assess how to rate the Trench generation 3 part using the newrules with a more accurate reflection of its true performance. Figure 1 shows
the new line in comparison with the two data sheet lines.

340

10
(1)
(2)
(3)

Zth (j-mb)
(K/W)
1

10-1

10-2
10-6

10-5

10-4

10-3

(1) BUK9Y38-100E(2) BUK9Y30-75B(3) BUK9Y30-75B (new test method)

10-2

10-1
1
tp (s)

aaa-013060

Figure 1 | Comparison of transient thermal impedance

Although there is a difference in the Rth, it is probably unimportant. In practice, it is
the Rth(j-amb) that is the limiting factor for the design. The Rth of the Printed-Circuit
Board (PCB) that is common to both parts, is dominant.
When considering the old test method with the new one for the BUK9Y30-75B, theother region of difference is below 10 μs.
For pulse duration between 1 μs and 2 μs, the temperature rise (or Zth(j-mb)) in the
Trench generation 3 part, is only a half of what the original data sheet curve
predicted. The importance of this factor depends on the application.

3.2 Q: It is understood that the values forthermal resistance listed in data sheets are
based on controlled conditions that do not
apply to typical applications. If thisunderstanding is true, how is the proper
thermal resistance/junction temperatureaccurately calculated?

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MOSFET & GaN FET Application Handbook

A: This understanding is correct. To ensure reliability of the MOSFET, always limit
the maximum junction temperature to 175 °C.

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Additional information
It is understood that the typical values for thermal resistance listed indata sheets are based on controlled conditions that do not apply totypical applications.
Device characterization at a junction temperature of 25 °C is the accepted
standard in the semiconductor industry. It is also most convenient for users totake measurements at this temperature.
How is the proper thermal resistance calculated?
Only a maximum value of thermal resistance is given on Nexperia MOSFET data
sheets. The typical value is significantly less than the maximum. It is understoodthat thermal cycling can induce an increase in Rth(j-mb) over the lifetime of the
MOSFET.
A tolerance margin is included in the data sheet maximum Rth(j-mb) value whichallows for an increase over the lifetime of the MOSFET.
For worst case design analysis, always use the maximum value. Maximum
Rth(j-mb) given on the data sheet is evaluated from characterization
measurements.
Its value is not dependent on temperature or other environmental conditions.
How is junction temperature calculated?
MOSFETs usually operate with a junction temperature greater than 25 °C due totemperature rise caused by the environment and/or power dissipation in the
MOSFET.
If MOSFET power dissipation and mounting base temperature (Tmb) are known,
MOSFET junction temperature can be calculated. Use Equation (1), below todetermine Tj.
Tj = P × Rth(j-mb) + Tmb					[1]SPICE thermal models of MOSFETs provide an excellent means of estimating Tj by
simulation. It is particularly useful when MOSFET power dissipation changes withtime.
Worked example for a BUK7Y12-40E:
From the data sheet:
Maximum RDS(on) at 25 °C = 12 mΩ

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Maximum RDS(on) at 175 °C = 23.6 mΩ
Maximum Rth(j-mb) at 2.31 K/W
From the application data:
PWM frequency = 100 Hz
Maximum duty cycle = 50 %Vsupply = 14 V
Rload = 0.7 Ω
Maximum ambient temperature = 85 °C
Maximum PCB temperature = 100 °C
Calculation based on average power, ignoring any temperature fluctuation
due to the power pulsing and also ignoring switching losses at 100 Hz:
Assume that the temperature of the MOSFET is initially 100 °C and its
maximum RDS(on) is 18 mΩ. It is midway between 12 mΩ at 25 °C and 24 mΩ at
175 °C.
When conducting, the MOSFETs power dissipation I2 × RDS(on) is: 20 x 20 x
0.018 = 7.2 W
The duty cycle is 50 %, so the average power dissipation = 7.2 x 0.5 = 3.6 W. It
is assumed that the switching loss at 100 Hz can be ignored.
The rise of the MOSFET junction temperature, above the mounting base is:2.31 x 3.6 = 8.3 K.
The maximum MOSFET die temperature in this situation is very safe at: 100 +8.3 = 108.3 °C
To guarantee that the PCB temperature does not rise above 100 °C in an 85 °C
ambient, the thermal resistance between PCB and ambient must be: (100
- 85)/3.6 = 4.2 K/W

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3.3 Q: What effect does changing the device
have on Rth(j-a)? The customer is trying toachieve ~60 K/W on a dual N channel FET.
A: The customer is trying to achieve a Rth(j-amb) = 60 K/W using a dual N channel
LFPAK56 (SOT1205).

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Nexperia | A Power Design Engineer's GuideRth(j-amb) = Rth(j-amb) + Rth(mb-amb), see Figure 2.

Tmb
Tj

Tamb
Rth(j-mb)

Rth(mb-amb)

Figure 2 | MOSFET thermal resistance junction to ambient

Rth(j-mb) is in Nexperia’s control (it is a function of the die size and package design,for example the bigger the die the lower the Rth(j-mb) ). Rth(mb-amb) is a function of the
PCB design and the thermal management scheme and is not under the control of
Nexperia. A very good multilayer FR4 design with thermal vias would be around 30- 40 K/W.
Worked example for BUK9K6R2-40E and BUK761R6-40E
From the BUK9K6R2-40E data sheet, the Rth(j-mb) is 2.21 K/W.
However, the Rth(mb-amb) is in the control of the customer, i.e. it depends on the PCB
design. Assume the PCB assembly thermal resistance to ambient is 40 K/W.
A. Using a SOT1205 dual LFPAK56 (such as BUK9K6R2-40E)
Rth(j-amb) = Rth(j-mb) + Rth(mb-amb) = 2.21 + 40 = 42.21 K/W
B. Using a device with a larger die will result in a lower Rth(j-mb). For demonstration
purposes a max die D2PAK (BUK761R6-40E) with low Rth(j-mb) is used (0.43 K/W).
Rth(j-amb) = Rth(j-mb) + Rth(mb-amb) = 0.43 + 40 = 40.43 K/W
You can see that for DC conditions, the effects of changing the die size on thedevice is swamped by the PCB cooling arrangement, Rth(mb-amb), (note that the Rth(j-a)
of 50 K/W in the data sheet is for a different PCB thermal arrangement).
So transposing this into temperature: ΔT = P x Rth(j-amb), then for a dissipation of 1
W:
A. 42.21 K/W: if Tamb = 105 °C then Tj = 105 + 42.21 =147.21 °C
B. 40.43 K/W: if Tamb = 105 °C then Tj = 105 + 40.43 = 145.43 °C
Rth(mb-amb) will change due to the package size, this effect is ignored here. The
bigger package will result in a lower mounting base to ambient thermal resistance.

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In example B above, Tj will probably be a few degrees lower than calculated. Refer
to AN90003 (chapter 9 in this book) - LFPAK MOSFET thermal design guide formore detailed information.
To estimate Rth(j-amb), bias the body diode of the MOSFET with 1 W on the intended
PCB design. Use an infra-red camera or sensor to measure the temperature of the
MOSFET plastic body, this is approximately the same temperature as the junction.

3.4 Q: Why is there no Rth(j-amb) in the data
sheet?
A: The Rth(j-amb) is dependent on the PCB design. As MOSFET manufacturers we do
not determine this part of the system and the value would be meaningless,therefore. We have provided some examples in (chapter 9 in this book) - LFPAKMOSFET thermal design guide for more detailed information.

3.5 Q: How can I achieve the Rth(j-mb) – I try to
put the rated power in the device, but thejunction is getting too hot?
A: Rth(j-mb) tells you the temperature difference between the junction and mounting
base for a given power profile. Because of the power dissipation the mounting
base to ambient path will also heat up, causing the junction temperature to risefurther. The junction to ambient is the full thermal path that needs to beconsidered and is a function of the PCB design too, please see AN90003 for moredetails. (chapter 9 in this book) - LFPAK MOSFET thermal design guide for moredetailed information.

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3.6 Q: Why is there no Rth(j-case) in the data
sheet?
A: The drain tab (mounting base) and source leads are the two main paths throughwhich a down side cooling package dissipates heat. In fact, contrarily to somethrough hole packages (like TO-220), SMD packages such as LFPAK and D2PAK get
rid of all the heat through the PCB. Hot air rises from the board and envelopes thedevice lowering the efficiency and thus the efficacy of any heatsink attached to the
top of the plastic case. Instead, when a substantial power needs to be dissipated,copper traces, vias and planes are employed in order to lower as much as possible
the Rth(j-a) of a device.

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3.7 Q: Can I measure Tj from Tcase?
A: Flotherm simulations and measurements carried out using LFPAK56 and variable
power dissipation and PCB copper area show how, in steady state conditions,temperature taken on the top center of the case is, within a reasonably low
accuracy, very similar to the junction temperature. This result is not due to heatbeing dissipated from the top of the case but rather from the one coming out of
the PCB that increases the temperature of the surrounding air immediately close to
the device, up to almost that of the junction.

3.8 Q: Why can an LFPAK88 have higher Rth(j-mb)
than an LFPAK56E?
A: Conduction is the predominant phenomena regulating heat flow from junction
to mounting base. The resulting resistance is inversely proportional to the crosssectional area of the medium through which it propagates (die area) and directly to
its thickness (drain tab). Given an LFPAK56E and an LFPAK88 with the same die size
the former has lower Rth(j-mb) because the thickness of its drain tab is lower. It is
worth noting, however, that the thermal path doesn't end here and that the
LFPAK88 shows better thermal performances due to its lower Rth(j-amb) given by amuch larger drain tab.

3.9 Q: Does the LFPAK88 have better or worse
thermal impedance than LFPAK56?
A: For a given die size the LFPAK88 shows an overall better transient thermal
impedance Zth(j-mb).
Additional information
Referring to Figure 3 we can see a much lower impedance in the region 20 μs to
10 ms. This is due to the larger mounting base of the LFPAK88. The greater
volume of copper acts as a heatsink with larger thermal capacitance that allows
a faster absorption of heat from the die into the drain tab itself. This isbeneficial in short circuit events, inrush current events as well as other highcurrent events.
During steady state the LFPAK56E shows lower thermal resistance Rth(j-mb) due
to the thinner drain tab (see Section 3.8).

346

1
Zth
(K/W)

BUK7J1R0-40H(LFPAK56E)

10-1

BUK7S1R0-40H(LFPAK88)

10-2

10-3

10-4
10-7

10-6

10-5

10-4

10-3

10-2

10-1

Based on thermal simulation

tp (s)

1

aaa-031684

Figure 3 | LFPAK88 (BUK7S1R0-40H) and LFPAK56E (BUK7J1R0-40H) transient thermal impedancecomparison

4 MOSFET body diode4.1 Q: How much current can the MOSFET body
diode carry?
A: The data sheet states the IS capability for the diode. The power constraints are
the same as for the MOSFET conduction. The diode is an integral part of the
MOSFET structure. They are in effect the same size and have the same thermal
properties. The MOSFET can carry the same current through the channel or inreverse through the body diode. The maximum steady state current in the diode isdependent on the total allowed power loss for the device. However, the diodecurrent may be different from the channel current because the power dissipation
may be different under the 2 modes of operation.

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MOSFET & GaN FET Application Handbook

Additional information
The objective is to keep the junction temperature below Tj(max) so it is necessary
to calculate the diode dissipation. In a DC case, it is simply IF x VF. It is equivalent
to VSD x IS as used in the Nexperia MOSFET data sheet. For a worse caseanalysis, the max VF of the data sheet should be used (normally 1.2 V).
Where the current is varying but in cyclic manner, the dissipation can be foundusing Equation (2) below:
Power = VO × I(AV)+ RS × I2 (RMS)

			

[2]

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where:
I(AV) is the average diode current
I(RMS) is the RMS diode current
RS is the slope of the ISD/VSD characteristic graph given in data
VO is typically where the RS line meets the axis at ISD = 0. For a conservative worst
case analysis, use 1.2 V.
100
ISD
(A)
80

60

40

20
(1)

0

0

0.5

(2)

(3)

1.0

(1) characteristic at 25 °C(2) idealized characteristic at 25 °C(3) worst case idealized characteristic at 25 °C

VSD (V)

1.5

aaa-013061

Figure 4 | Diode characteristics

For transient currents, a simulation using the SPICE model of the diode is useful but
care is needed because the model is for a typical part. Once the dissipation is
known, standard thermal analysis methods can be used to check that Tj is
acceptable.
It can include SPICE simulation using RC thermal models.

4.2 Q: What is the importance of the Qr
parameter?
A: When a MOSFET transitions from diode conduction to blocking state there is anadditional loss, called the diode recovery charge (Qr). The Qr needs to be factored
in the switching loss calculation of the application for accurate analysis. This
switching transition also impacts on the EMC performance and needs carefulconsideration - see AN90011 (chapter 7 in this book) - Half-bridge MOSFET

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switching and its impact on EMC and TN90003 (chapter 9 in this book) - LFPAKMOSFET thermal design for more details.
Additional information
Qr is relevant in all half-bridge topologies, for example DC-to-DC convertors or
motor control. In both examples the MOSFET's body diode will be conducting in
the forward direction before it is required to block voltage in the reverse
direction. This change of operation is called diode recovery and takes time
(reverse recovery time, trr) during which charge is stored (diode recovery
charge, Qr). The Qr adds additional losses to the switching transient.
In addition to switching losses the recovery of the diode is related to the EMC
performance of a MOSFET in an application. A fast recovery will cause highvoltage spikes and a poorly damped diode will cause voltage and current
oscillations after the switching event causing EMI.

5 Safe operating area and linear mode
operation
5.1 Q: The current derates with temperature.
Is this limit based on power dissipation?
A: The most important factor in current derating or power derating is junction
temperature. Tj is a function of power dissipation. Power dissipation is a function of
ID current and on-state resistance (P = ID2 × R) when operating in the fully enhanced
mode. It is the product of ID and VDS when operating between on and off states.
The RDS(on) of a MOSFET, increases with increase in temperature. Therefore, for agiven maximum power dissipation, the maximum current must be derated tomatch the maximum power dissipation. In Nexperia data sheets, graphs show thecontinuous drain current and normalized total power dissipation (see Figure 6) as afunction of the mounting base temperature. These graphs can be used todetermine the derating.

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5.2 Q: Is it necessary to de-rate any limit(current, voltage, power etc.) to achieve highreliability?
A: If current, voltage, power, junction temperature, etc. are within Nexperia data
sheet limitations, no additional derating is needed. In the data sheet, there is a349

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power derating curve based on junction temperature. Junction temperature (Tj) is
one of the most important factors for reliability. Particular care should be taken to
extract enough heat from the device to maintain junction or die temperature,
below rated values. The device should be operated within the SOA region. It should
be de-rated if necessary (see Section 5.3) as recommended in the data sheet and it
should be possible to obtain optimum reliability.

5.3 Q: How do I de-rate an SOA graph fortemperatures other than 25 °C?
A: As an example, assume that the temperature required is 100 °C, instead of 25 °C.
Tj rated is 175 °C for this automotive grade MOSFET. To de-rate when considering
the effect of temperature on SOA performance the current must be reduced, (seeFigure 5). To determine the new current (at temperature) for a fixed voltage, use
the power derating line in Figure 6. For example, power at 100 °C = 50 % of power
at 25 °C. Therefore, the 1.0 A line represents 0.5 A at 100 °C etc. It is explained in
Application note AN11158 (chapter 1 in this book) - Understanding power MOSFET
data sheet parameters. If necessary, the SOA lines for 1 ms, 10 ms etc. can beextended at the same slope to the right.
ID(A)

30 A at 100 °C

103
Limit R DS(on) = VDS / ID
102

tp = 10 us

10DC
0.5 A at 100 °C

1 ms

1

10 ms
100 ms0.05 A at 100 °C

10-1
10-1

1

10

Tmb = 25 °C; IDM is single pulse

Figure 5 | Example: BUK7Y12-55B SOA curve showing derating for 100 °C

350

VDS (V)102

aaa-002475

120
Pder
(%)
80

40

0

0

50

100

150

Tmb (°C)

200

03aa16

Figure 6 | Normalized total power dissipation as a function of mounting base temperature

Ptot
x 100%
Pder = ______
Ptot (25ºC)

				

[3]

5.4 Q: Is there a Spirito boundary limit line forlinear mode operation?
A: The Spirito region or hot spotting issue with new higher density technologies
may have more effect in the linear mode of operation. This effect is evident fromthe change in gradient in the limit lines for 1 ms, 10 ms and 100 ms at higher VDS
values (see Figure 5). The 1 ms, 10 ms, 100 ms and DC lines at higher VDS values
emphasize it. The reason is that most newer technologies pack more parallel
fundamental cells to share more current in a smaller die (lower RDS(on) per unit
area). It leads to an increased thermal coupling between cells. Also, to attain higher
current densities, the MOSFETs are designed with higher transconductance or gain
(gfs = ID/VGS). It enables them to carry higher currents even at lower VGS values.
However VGS(th) (threshold voltage) has a negative temperature coefficient whichleads to a higher zero temperature coefficient crossover value. For various reasons,the distribution of temperature in the die is never perfectly uniform. Therefore,
when the device is operated for extended periods in linear mode, hot spotting
occurs. Due to the shift in threshold voltage, there is a risk of thermal runaway anddevice destruction where the hotspots form. Because of these reasons, special careshould be taken when using trench or planar MOSFETs for linear applications.
Ensure that operation remains within the data sheet SOA limits.

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5.5 Q: Regarding the SOA curve for the 1 ms
and 10 ms curves, there is a bend-down at
higher voltages and low currents. Why does
this bending disappear for longer and shorter
pulse times?
103
ID
(A)
102

Limit RDS(on) = VDS / ID
tp = 10 µs
100 µs

10
DC
1

1 ms
10 ms
100 ms

10-1
10-1

1

10

102

VDS (V)103

aaa-022335

Figure 7 | Safe operating area: continuous and peak drain currents as a function of drain-source
voltage

A: The inflexion points on the 1 ms and 10 ms lines represent the points where the‘Spirito’ effect starts. At higher ID, the lines represent constant power (P); at lower
ID, P decreases as ID decreases. The 100 ms and DC lines are straight, but havehigher negative gradients than constant power lines, i.e. power also decreases as ID
decreases. The flat portion of the DC line represents package maximum ID.
The Spirito effect is a form of electro-thermal instability i.e. uneven die heating
leading to hot-spot formation. It happens because VGS(th) has a Negative
Temperature Coefficient (NTC) at ID values below IZTC (zero temperature coefficientcurrent). The consequence is to reduce MOSFET power dissipation capability inlower ID zones of the SOA chart.
Measurement at DC, 100 ms, 10 ms and 1 ms establishes SOA capability. The 100 μs
and 10 μs lines on this graph are theoretical constant power lines. They are realistic,
as the Spirito effect is much less significant at higher currents and shorter pulse
periods.
Reliable 100 μs SOA measurement capability has recently been achieved, so futuredata sheets include 100 μs SOA lines based on measured data. It is now evident
that the Spirito effect is apparent at 100 μs. Consequently, from 2016, new352

MOSFET releases have a measured 100 μs SOA line in their data sheet SOA graph.
See AN11158 (chapter 1 in this book) - Understanding power MOSFET data sheet
parameters for further information.

5.6 Q: How does Nexperia ensure compliancewith the SOA curve during series production?
A: The factors influencing the compliance of the MOSFET with the data sheet SOAgraph are:• the uniformity of the MOSFET cells across the active (trench) surface of the die• the integrity and uniformity of the die attachment (the solder layer between thedie bottom (drain) surface and mounting base)
Cell uniformity must be good for the MOSFET to work. However, cell uniformity cannever be perfect and there is always some variation between cells.
The integrity of the soldering to attach the die must be good without voids or die
tilt. If not, the local (junction to mounting base) thermal impedance varies withlocation across the die. It gives uneven cooling. Uneven die surface cooling may be
due to either or both of the factors stated. However, the consequence is the samei.e. SOA non-compliance with the data sheet graph.
In production, linear mode power pulse tests are used to stress the MOSFET
thermally. If the die cooling is not sufficiently uniform, hotspots can form and thedevice parameters can change more than expected. A decision to reject parts canbe made based on the results.

13Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

5.7 Q: How can a part be identified when it isdesigned for linear mode operation?
A: While all Nexperia MOSFETs can be used in linear mode operation, someNexperia MOSFETs are designed specifically to be used in linear mode. The device
description in the data sheet states that the device is suitable for operation in linear
mode. To determine the suitability for operation in linear mode, perform athorough analysis of the SOA graph. This analysis includes derating the SOA graphfor junction temperatures above 25 °C. The naming convention indicates that the
MOSFET is designed for linear mode applications.

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5.8 Q: For parts designed for linear mode
operation, are there any restrictions (such as
the Spirito boundary)?
A: Even if a MOSFET is intended for use in linear mode applications, the part
must not be operated outside its SOA. Post 2010, all Nexperia MOSFETs have ameasured SOA characteristic. The limit of linear mode capability on Nexperia parts
is shown in the SOA characteristic. As a result, the boundary of what is safe isestablished via measurement rather than calculation. The Spirito capability limit isshown in the SOA characteristic.

6 Avalanche ruggedness and Unclamped
Inductive Switching (UIS)
6.1 Q: Are trench designs susceptible to the
UIS issue (parasitic BJT turn-on)?
A: In general - Yes, but Nexperia Trench MOSFETs are designed to suppress thiseffect. The trench structure, unlike planar, can be very easily designed to suppress
parasitic turn on of the BJT. For new Nexperia MOSFET technologies, the failure
mechanism is thermal, which represents the limit of achievable UIS performance.
Planar (on the left in the diagram) and trench (on the right in the diagram) MOSFET
technologies, are shown in Figure 8. In the trench case, a design feature in thesource contact effectively short circuits the base-emitter of the parasitic BJT. In
older planar technology, the shorting of base to emitter of the parasitic bipolar is
not as effective. It is due to the longer path length in the n and p regions.
S

G

S

n

n

G

n

D

Figure 8 | Simplified planar and trench technology

S

n

n

p

p

354

S

p

p

n

n

D

aaa-015947

6.2 Q: Why are planar designs susceptible tofailure during UIS?
A: All MOSFETs are susceptible to failure during UIS. It depends on whether the
MOSFET Tj reaches the intrinsic temperature of silicon. Furthermore, if the parasitic
BJT is triggered, they can fail even earlier. It is because the BJT can be switched onrelatively quickly but is slow to switch off. Current can then crowd in a certain partof the device and failure results. Newer Nexperia trench technologies are lessvulnerable to triggering of the BJT than planar designs. See Section 7.1 and the
additional information associated with Section 6.3.

6.3 Q: How are parts constructed to minimize
failure during UIS?
A: The base emitter path in the silicon design is designed to minimize the risk oftriggering the parasitic BJT.
Additional information
There are two strategies employed to prevent triggering of the parasitic BJT:
1. Reduce the avalanche current per cell - To reduce the avalanche current per cell,
there must be a higher cell density. It is easily achieved with trench technologies
but more difficult with planar devices. If improvements in planar technologies
are seen, it is likely that modern fabrication equipment achieves a high enoughcell density. Increasing cell density too much, deteriorates linear mode 		
performance (Spirito boundary occurs sooner), so cell density is a trade-off. For
this reason, Nexperia has not been as aggressive as other trench MOSFET 		
manufacturers in achieving very high cell densities. The target is to ensure thatthe thermal limit in UIS is achieved.

13Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

2. Ensure that the current flow during avalanche does not flow through the 		base-emitter region of the BJT. This feature is the significant advantage oftrench over planar. The parasitic bipolar is formed between the source of the
MOSFET, the body region (i.e. channel) and the epi region (i.e. the drain). If there
is enough avalanche current through body junction, there may be enoughvoltage developed to bias on the BJT leading to device destruction. In Nexperia
devices, a modified source contact is used. This contact collects any avalanche
current, preventing it from biasing the BJT on.
For planar devices, strategies include reducing the gain of the BJT by placing
high doped implants close to the channel. It is similar to (2) in intention but it is
not as effective.

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6.4 Q: Is 100 % UIS testing required on
MOSFETs?
A: UIS testing is a fundamental part of Nexperia's defect screening procedures. It isapplied to all devices. The test is designed to increase the junction temperature to
Tj(max).

6.5 Q: I have parts not capable of parasitic BJT
turn-on. Why?
A: Devices fail at the thermal limit. At the thermal limit, the silicon becomes
intrinsic and blocking- junctions cease to exist. It is considered to be the only
UIS-related failure mechanism in our devices.

6.6 Q: What is the chart accuracy for avalanche
current versus time in avalanche, or energy
versus junction temperature?
A: Avalanche current versus time graphs are based on conditions that take a device
to Tj(max) and therefore, our ruggedness screening covers them. All Nexperia
MOSFETs are ruggedness tested during assembly and characterized during
development. The graphs are accurate and provide the worst case capability of thedevice to ensure reliability.

6.7 Q: For energy versus junction temperature
charts (if applicable), how is the inductance,
maximum current, time in avalanche etc.,
determined from the chart?
A: A temperature rise model is used, which is shown in AN10273 (chapter 4 in thisbook) - Power MOSFET single-shot and repetitive avalanche ruggedness rating.
Additional information
Although energy levels for UIS are often quoted on data sheets, a single
number can be misleading. Therefore a graph is provided, that outlines
conditions that take junction temperature to Tj(max). The user must determine
the current/time in avalanche based on the particular conditions. Examples are356

provided in AN10273 (chapter 4 in this book) - Power MOSFET single-shot andrepetitive avalanche ruggedness rating.

6.8 Q: Are repetitive avalanche ratings thesame as for a single pulse?
A: No. The repetitive avalanche ratings are lower than the single pulse rating. Referto the product data sheet for the device capability. An example is shown in
Figure 9.
Additional information
Repetitive means that the avalanche event is an intended operating condition
for the device.
Similarly, single-shot means that the MOSFET is expected to experience anavalanche event as a result of some unintended fault condition. Only 1 fault canoccur at a time, the MOSFET must cool to the starting temperature and thejunction temperature must not exceed 175 °C. Degradation of device
characteristics is likely after a relatively low number of occurrences.
Nexperia shows both single shot and repetitive avalanche capability in the
MOSFET data sheet. Generally, the repetitive current is 10 % of the single shotcurrent capability for a given inductor (so the time in avalanche is shorter, see
Figure 9 and Section 6.14).
For calculating repetitive avalanche ratings, calculate the starting junction
temperature for each avalanche incident independently. The calculations are
based on the frequency and duty cycle of avalanche condition and summed
over the entire period of expected repeated avalanche.

13Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

This topic is discussed in detail in Application note AN10273 - (chapter 4 in thisbook) - Power MOSFET single-shot and repetitive avalanche ruggedness rating.

6.9 Q: Are there any special failure modes
associated with repetitive avalanche?
A: The device can sustain small amounts of damage with each avalanche event andover time they can accumulate to cause significant parametric shifts or device
failure. Nexperia has performed research into this area and provides the repetitive
ratings in the data sheet. See also Application note (chapter 4 in this book) - PowerMOSFET single-shot and repetitive avalanche ruggedness.

357

Power and small-signal MOSFET frequently asked questions and answers

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6.10 Q: How does the increase in cell density
affect avalanche capability of MOSFETs?
A: There are two failure modes: current (parasitic BJT turn-on) and thermal. Celldensity has implications for these failure modes.
Additional information
Current - If enough avalanche current flows through a cell, a voltage drop
occurs in the p- region of the device as the avalanche current flows to thesource contact. This volt drop occurs in the base of a parasitic bipolar device. In
this mode, the resistance in the p-region/ base and the avalanche current (I =V/R) are important. Once VBE reaches the bipolar switch- on threshold, the
MOSFET is destroyed (VBE reduces with temperature). So at higher celldensities, for the same avalanche current, there are more cells and current percell is reduced. Each individual cell has less current and is less likely to trigger
the parasitic device. It means that the total die current, required to cause adevice to fail, increases. Additionally, since the cell is smaller, the path throughthe p-region to the source contact is reduced. It makes it even harder to trigger
the parasitic device and again increases the current required to destroy thedevice.
Thermal - If the avalanche current is such that the parasitic BJT is not triggered,the device heats up. If the avalanche energy is sufficient, the silicon die reaches
temperatures at which the device starts to become intrinsic. The blockingjunction no longer exists, resulting in the destruction of the device. It is what ismeant when a reference is made regarding failure due to reaching the thermal
limit of a device. If the failure mode is thermal, changes in technology cannot
improve things significantly. New technologies are generally more robust inavalanche conditions. Note, if a thermal limit is reached, the only solution is toimprove the thermal impedance at the device level. Moving to a smaller die canbe detrimental.
Summary - New technologies improve the high current avalanche capability ofa device due to increased cell density and reduced parasitic NPN baseresistance. Lower current, higher energy (i.e. longer duration) avalanche
capability is unchanged.

6.11 Q: How many times can a device sustain
single avalanche events?

Example - A device has an avalanche event once in two months so how many cycles
of such an avalanche frequency can the device sustain? This question relates more
to quality and reliability but it is important nonetheless.

358

A: Refer to Section 6.8 of this chapter. For the answer to this question, refer to
Section 2.4.3 of AN11158 (chapter 1 in this book) - Understanding power MOSFET
data sheet parameters and all of AN10273 (chapter 4 in this book) - Power MOSFET
single-shot and repetitive avalanche ruggedness rating..
Additional information
Keep each avalanche event within the safe limits for repetitive operation
specified on the data sheet and Tj below 175 °C. There should be no
degradation of the MOSFET characteristics and no impact on MOSFET quality or
reliability. There are some applications where MOSFETs are repetitively
avalanched (e.g. some engine controllers) and the reliability is good. Although
this condition takes VDS beyond the data sheet maximum, the data sheet also
specifies a maximum avalanche energy.
Extensive avalanche testing is performed on Nexperia MOSFETs. All theindications are that they are very robust. It is understood that most MOSFETs inautomotive applications are likely to experience avalanche conditions at somestage during their lifetime. It could be due to occasional fault conditions or as aconsequence of the circuit design (e.g. ABS solenoid valve driver MOSFETs).

6.12 Q: Is it possible for the avalanche current
on a device to exceed the package maximum
current but not the die maximum current?
A: The current specified in the avalanche graph should not be exceeded. It is
restricted to the DC rated current. The device factory test defines the limit which isguaranteed for the device.

13
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MOSFET & GaN FET Application Handbook

6.13 Q: How is the avalanche rating on the
body diode obtained (testing or modeling)?
If it is tested, how is it tested and what circuit
model is used?
A: The avalanche rating is modeled first and the results are then verified by testing
to destruction. The test circuit used is similar to the one defined in JESD24-5. For
SPICE modeling, the reverse diode characteristics can be defined and modeled. By
adding an RC thermal model of the Zth characteristic, it is possible to estimate the Tjof the device.

359

Power and small-signal MOSFET frequently asked questions and answers

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Nexperia | A Power Design Engineer's Guide

Additional information
The body diode of the MOSFET is not a single circuit element, but a structure
distributed throughout the MOSFET. There is a diode element associated witheach cell. In behavior terms, it can be represented as a single (Zener) diode inparallel with the single MOSFET (representing the sum of all the cells). The
design of the MOSFET determines the avalanche rating. Its representation in
the model is based on parameters measured during characterization testing.
The constraints are the same as for the MOSFET conduction. Diodes are an
integral part of the MOSFET structure. They are in effect the same size andhave the same thermal properties. The objective is to keep the junction
temperature below Tj(max) so it is necessary to calculate the diode dissipation.
For transient currents, the simulation using the SPICE model of the diode isuseful but care is needed because the model is for a typical part.
Once the dissipation is known, standard thermal analysis methods can be used
to ensure that Tj is acceptable.

6.14 Q: How is the repetitive avalanche safe
operating area derived in the data sheet
graph? The repetitive avalanche SOA curve
seems to be the same as single-shot Tj = 170 °C.
A: The repetitive line is the line for a start temperature of 170 °C. It is because it
predicts a temperature rise of 5 °C which is the maximum permissible rise from any
starting temperature, see AN10273 (chapter 4 in this book) - Power MOSFET
single-shot and repetitive avalanche ruggedness rating.). It also corresponds to
10 % of the single-shot current using the same inductor value, see Figure 9.
Additional information
The reason it applies to any temperature is because the temperature does notstrongly influence the wear-out caused by repetitive avalanche. The strongest
influence is the current.
The avalanche current is composed of high energy charge carriers moving
through the depletion region. As they pass through, they can collide with the Si
structure. There is a chance that a high energy carrier (sometimes called a hot
carrier) is produced that collides with the gate oxide causing damage. It is notcompletely destroyed but it does cause it to wear out, which is observed asparameter variation.

360

Higher currents mean more electrons, more collisions and more frequentdamaging events leading to faster wear-out and lower reliability. The target
should be a failure level <0.1 ppm over the full vehicle life. Experiments indicatethat if repetitive current is limited to 10 % of single-shot current capability for
Tj(start) = 25 °C, it results in a Tj increase of 5 °C. There is no significant
degradation of the device.
For example: A 15 mH inductor carrying ~19 A gives an avalanche time of ~5
ms. It has a peak temperature rise of 150 °C putting it on the limit line for Tj(start)= 25 °C. By reducing the current to 10 % = 1.9 A, tAL reduces to ~500 μs and thetemperature rise is 5 °C.
103
IAL
(A)
(1)

102

(2)

10
(3)

1
(4)

10-1
10-3

10-2

10-1

(1) = characteristic at 25 °C(2) = characteristic at 150 °C(3) = repetitive characteristic at <170 °C(4) = inductor characteristic

Figure 9 | Avalanche safe operating area

1

tAL (ms)10

aaa-013064

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7 Capacitive dV/dt issues
7.1 Q: Is there a particular capacitance or
charge ratio that should be used to prevent
turn-on, or is it circuit dependent?
A: The capacitive dV/dt turn-on is strongly circuit dependent.
If the dV/dt across the MOSFETs drain to source is too high, it may charge CGD,
which is the capacitance between drain and gate, inducing a voltage at thegate. The gate voltage depends on the pull-down resistor of the driver basedon Equation (4):
VGS = Rdriver × CGD × dV/dt 					

[4]In some bipolar drive circuits, such as emitter follower derived circuits, theproblem is increased. It is because the driver cannot pull the gate down to 0 V
and has approximately 0.7 V offset.
It is also important that the driver is referenced to the MOSFET source and not
to signal ground, which can be significantly different in voltage.
The ratio of CGD to CGS is a factor but a good drive circuit is the critical factor.
Even if a VGS spike is present, it is safe for the MOSFET as long as the dissipation
is within thermal limits and MOSFET SOA limits.

7.2 Q: How are parts constructed to minimize
this effect?
A: Nexperia MOSFETs are designed with a high threshold at high temperatures andwe check VGS threshold at 25 °C is within data sheet limits. Logic level devices aredesigned and guaranteed to have a minimum threshold voltage >0.5 V even at
175 °C.
Additional information
Maintain a reasonable ratio between CGD and CGS. The gate network structureof the device is designed to have good control of all areas of the die.

362

7.3 Q: How is dV/dt characterized?
A: It is usually measured in a half-bridge test circuit. It is a measure of the device
dV/dt during body diode reverse recovery. This data is not normally published in thedata sheet. This dV/dt is in practice the highest dV/dt the device experiences.
Additional information
Failure due to dV/dt is not something seen in modern low-voltage MOSFETs
however, dV/ dt is normally characterized for Nexperia MOSFETs. The failuremode is that the capacitive current resulting from dV/dt, triggers the parasitic
BJT. However, as the voltages are low (dV/dt is more an issue > 600 V) a
current/charge high enough to trigger the parasitic BJT cannot be generated.

7.4 Q: What diode or other parameters areimportant to assess susceptibility?
For example, maximum dV/dt and maximum IF.
A: High dV/dt can induce glitches onto the gate of the MOSFET. A snubber can help
to reduce dV/dt and the magnitude of the VDS spike if significant. The ratio of Coss at
low VDS compared to Coss value at high VDS is an indicator of the non-linearity of Coss.
A very high ratio can indicate that the device can generate a high dV/dt. Gate drivercircuit design can reduce the gate glitch, see Section 7.1. The ratio of QGD to QGS
and the gate threshold voltage can be used to indicate the susceptibility of thedevice to gate glitches.

7.5 Q: Is trench technology sensitive to thisphenomenon?

13
Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

A: In theory, all MOSFETs are.
Additional information
dV/dt induced turn-on of the parasitic bipolar transistor is not known as an issue
in low voltage Nexperia MOSFETs. If UIS parasitic turn-on is solved, then dV/dt
induced turn-on is also solved. Refer to Section 6.1 and Section 6.3 for more
information.

363

Power and small-signal MOSFET frequently asked questions and answers

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Nexperia | A Power Design Engineer's Guide

7.6 Q: Does a soft recovery body diode givelower dV/dt and if so, how is it designed andfabricated into the part?
A: Soft recovery does reduce the dV/dt. Although dV/dt is not an issue for the
MOSFET, a lower dV/dt is better for EMI, voltage spikes and crosstalk. The design
and manufacture is very specialized, involving proprietary information.

7.7 Q: How does temperature influence thissensitivity to dV/dt and why?
A: At high temperatures, it is easier to trigger a parasitic bipolar as its VBE reduces.
But if the BJT is effectively shorted out and current diverted away from it, as
discussed in Section 6.1, then it is not an issue.

7.8 Q: Can Nexperia provide RB, CDB, VBE
saturation values in the parasitic BJT model, as
shown in Figure 11?
A: These values are required to be able to calculate Equation 6. The aim is to obtaina dV/dt value to check if parasitic BJT turns on, leading to device failure. It is
impossible to measure the characteristics of the parasitic bipolar transistor as its
terminals cannot be accessed independently of the MOSFET terminals. A parasitic
bipolar transistor is always created when a MOSFET is fabricated. Referring to
Figure 10 it can be seen that there are two current paths which could cause
MOSFET problems. Current I1 flows via CGD and depending on CGS and RG it cancause the MOSFET to switch on momentarily. It is often referred to as a gate glitch.
Current I2 flows via CDB and RB which can potentially switch on the parasitic BJT.
Additional information

364

VGS = I1 × RG = RG × CGD × dV/dt				[5]

VBE = I2 × RB = RB × CDB × dV/dt				[6]

DRAIN

D
I1

CGD

CDB
NPN
bipolartransistor

G
I2
CGS

RG

appliedramp
voltage

RB

S

SOURCE

aaa-013073

Figure 10 | Power MOSFET showing two possible mechanisms for dV/dt induced turn-on

SOURCE

GATE

N+
RB

CDB

P

N

DRAIN

aaa-013074

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

Figure 11 | Physical origin of the parasitic BJT components that may cause dV/dt induced turn-on

Additional information
Early lateral MOSFETs (structure shown in Figure 11), were susceptible tofailure caused by the turn-on of this parasitic transistor.
Modern power MOSFETs have greater cell density meaning the current per cellis reduce and the voltage across RB is lowered. In addition, the geometry ofeach individual cell is smaller compared to previous generations making RB
shorter in length and therefore a lower value of resistance. Both these greatly
improving the immunity of modern trench MOSFETs (structure shown in Fig.
12) to dV/dt latch-up.

365Nexperia | A Power Design Engineer's Guide

In application tests we see dV/dt values in the order of 1 – 10 V/ns for T6 and T9technology when switched quickly using a high current gate driver with low
gate resistance.

Source

Oxide
N

Gate

Power and small-signal MOSFET frequently asked questions and answers

13

PN
Drift Region

Substrate
Drain

aaa-013075

Figure 12 | Trench generation 6 structure

7.9 Q: In a half-bridge configuration one
MOSFET is driven off, and the second MOSFET
begins to turn on. The first MOSFET thatshould be off is turning back on again causing
cross conduction and an over-current peak.
What could be the cause? See Figure 13 forphenomena.
12

6current (A)VGS (V)

overcurrent
peak
10

5

8

4

6

3

voltage(V)

MOSFET turning
back on4

20current
(A)

2

1

time

0

aaa-022330

Figure 13 | Blue VGS trace shows gate bounce and the red current waveform spiking when gate
bounce occurs

366

A: It is sometimes referred to as gate bounce. MOSFETs have internal stray
capacitances coupling all three terminals and the gate is floating. The capacitors
are inherent to the internal structure of a MOSFET, see Figure 14.

CGD
NMOS
CDS
CGS

aaa-022333

Figure 14 | MOSFET Schematic representing the internal capacitances

CGD and CGS form a capacitive potential divider. When a voltage appears across thedrain and source of the MOSFET, it couples to the gate and causes the internal gate
source capacitor to charge. If the voltage on the gate increases beyond the
MOSFET's threshold voltage, it starts to turn back on which can cause crossconduction. The ratio of the capacitances CGD and CGS determines the severity ofthis effect.
Additional information
Figure 15 shows a simplified circuit that can exhibit this behavior. It could be asynchronous buck regulator or one leg of a 3-phase inverter. In both cases, theload is inductive. A current source represents the inductive element in thiscircuit.

high-side
FETPower and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

low-sideFET

aaa-022334

Figure 15 | Simplified schematic of a MOSFET half-bridge. The high-side and low-side MOSFETs are
switched antiphase to each other367

Power and small-signal MOSFET frequently asked questions and answers

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Nexperia | A Power Design Engineer's Guide

17
voltage(V)
14

6
VGS low-side (V)VGS high-side (V)VDS low-side (V)Current (A)

11

4

8

3

5

2

2

1

-1

current
(A)

5

time

0

aaa-022331

Figure 16 | Waveforms showing the full detail of the gate bounce phenomenon

The waveforms in Figure 16 show the critical part of the applications operation
when the current commutates from one MOSFET to the other. In this instance, thehigh-side MOSFET is being switched on as the low-side one switches off. Current
flows from the switch node. Note the low- side MOSFET behaves as a diode when itis switched off. This behavior is similar when current flows into the switch node. In
this case, the high-side device becomes the synchronous rectifier. Referencing the
waveforms in Figure 16, the blue low-side VGS trace begins to drop and the lowside MOSFET turns off. The green low-side VDS trace going negative indicates that it
has switched off. The low-side MOSFET body diode begins to conduct the inductive
current and there is a VF drop of approximately 0.7 V. The high-side MOSFET begins
to turn on to a point where it conducts all of the load current. The low-side diodeturns off and the low-side VDS can now increase to the supply voltage. However, as
the VDS rises, the dV/dt is coupled back to the low-side gate through CGD. The blue
low-side VGS trace begins to increase again. In this example, the gate voltage rises
sufficiently above the threshold voltage so that it is now turned back on again.
Both high-side and low-side MOSFETs are on simultaneously. The current rapidly
increases due to the short circuit across the supply as indicated by the red current
trace.
There are several factors that influence this behavior, the most dominant of whichis the gate resistor. If the gate resistor is set to a level where the gate driver cannot
sink the capacitively coupled current, it must feed into the CGS. It causes the gate
source voltage to rise as can be seen in Figure 16.

368

12
voltage(V)
10

6
VGS at Rg = 100 Ω
VGS at Rg = 200 Ω
VGS at Rg = 300 Ω
Current at RG = 100 Ω
Current at RG = 200 Ω
Current at RG = 300 Ω

current
(A)

5

8

4

6

3

4

2

2

1

0

0

Each waveform has a different external gate resistor. Increasing the gate resistor has adetrimental effect on the severity of the gate bounce.

aaa-022332

Figure 17 | Effect of RG on VGS, and shoot-through current

It is good practice to keep the external gate resistor as low as possible. In reality,
there is a limit to how low this value can be, due to EMC considerations. Good
design practices can be employed to improve EMC while maintaining a low enoughvalue of gate resistor to prevent gate bounce. Simple alternatives to increasing thegate resistor include adding extra capacitors external to the MOSFET between gate
and source, located close to the MOSFET connections. It helps with radiatedemissions without affecting the gate turn off waveforms as much.
Other key factors include the MOSFETs threshold voltage (Vth). The lower Vth is, themore enhanced a MOSFET is for a given gate bounce voltage. It therefore increases
the severity of the cross conduction. Vth is also temperature dependent. It reduces
as die temperature increases, further compounding the severity.

7.10 Q: Does the BUK9Y29-40E fulfill thefollowing requirements?

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

Table 1: Turn on/off delay, rise time and fall time requirements
Symbol Parameter

Conditions

Min

Typ

Max Unittd(on) + tr

turn on delay time +rise time combined

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; R = 50 ;8
RG(ext) = 5 Ω

13

18

ns

td(off) + tf

turn off delay time +fall time combined

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; RL = 50 Ω;
9
RG(ext) = 5 Ω

14

19

ns

369

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

A: Typical values of td(on), tr, td(off), and tf are given in the BUK9Y29-40E data sheet:

Table 2: Turn on/off delays, rise time and fall time BUK9Y29-40E
specifications
Symbol Parameter

Conditions

Min

Typ

Max Unittd(on)

turn on delay time

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; RL = 5 Ω;RG(ext) = 5 Ω

6

-

ns

tr

rise time

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; RL = 5 Ω;RG(ext) = 5 Ω

7

-

ns

td(off)turn off delay time

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; RL = 5 Ω;RG(ext) = 5 Ω

9

-

ns

tf

fall time

Tamb = 25 °C; VDS = 30 V;
VGS = 5 V; RL = 5 Ω;RG(ext) = 5 Ω

5

-

ns

td(on) + tr must be between 8 ns and 18 ns. For the BUK9Y29-40E, this value istypically 13 ns.
td(off) + tf must be between 9 ns and 19 ns. For the BUK7Y29-40E, this value istypically 14 ns.
Minimum and maximum values for td(on), tr, td(off), and tf are not stated in Nexperia
MOSFET data sheets. It is because accurate and repeatable measurements of theseparameters, particularly tr and tf, very much depend on the test environment.
Even measurement probe positioning is critical because stray inductances,
capacitances and coupling fields change when the probe is moved (unavoidable).
A good worst case estimate tolerance on these parameters is ±50 %, however theseparameters have only a very small effect on MOSFET dynamic operation.
The typical BUK9Y29-40E fulfills these requirements, but if these parameters areslightly different, the consequences to the circuit switching performance are notsignificant. In the parametric test, a 50 Ω load resistor and 5 Ω external gate resistor
are used. It is very unlikely that these values are the same as the values used in theapplication.

370

Gate drive characteristics, MOSFET capacitances and the nature of the load circuit
have much more profound effects on the dynamic switching behavior of the
MOSFET. Specifically, its power loss, efficiency and EMC performance.

8 Package and mounting
8.1 Q: On the drawing for the power SO8/LFPAK56 common footprint, there are no vias on
the exposed pad. Are the addition of vias advised and, if so, which diameter?
A: If improved thermal resistance is required, vias can be added to the footprint.
The effect of adding vias is discussed in Section 3.5 of AN90003 (chapter 9 in thisbook) - LFPAK MOSFET thermal design guide.
Additional information
Nexperia has used 0.8 mm successfully but it does not mean that other sizes
would not work. The vias should be pre-filled with solder and hot air leveled, togive a flat surface, before the devices are placed.
If the vias are not pre-filled, there is a chance that the solder under the part isdrawn into the vias, which may result in voids.
The extra process steps on the PCB and the potential problems mean that vias
should not be used unless they are needed. Consult the manufacturing processengineers regarding surface mount and soldering process issues.

8.2 Q: How are devices tested for HV isolation
tests? An application is tested at
approximately 1 kV for HV isolation testing
across various terminals and a significant value
is seen across the MOSFET. Are there tests
that perform HV isolation analysis and, if so,
what are they?

13
Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

A: We do not perform any HV isolation tests on any automotive MOSFETs or specify
any HV isolation parameter in our data sheets. Insulation testing is only applicable
to TO-220F packages (Nexperia SOT186A)
Additional information
HV isolation is specified for MOSFETs with insulated drain tabs or in modules

371

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

with isolated bases. The test voltage applied in the Nexperia factory is 2.45 kV
for 0.4 seconds (Vrms at 50 Hz).

8.3 Q: The efficiency of my DC-to-DC converter
exceeds my requirements. Can I use smaller,higher RDS(on) MOSFETs to save money?
Environmental conditions: 4-layer FR4 board at 105 °C ambient temperature.
A: Although it is possible to reduce efficiency, other factors become theconstraints.
Additional information
The dominating factor is likely to be the temperature allowed at the solder joint
between the MOSFET and the PCB. It is unlikely that 125 °C may be exceededwith FR4. If the dissipation is 2 W, the thermal resistance of the path from the
MOSFET mounting base to ambient must be <10 K/W.
(125 °C - 105 °C)/2 W = 10 K/W.
Some special arrangements are required to achieve this figure. However thecustomer has indicated an allowed dissipation of 2 W so they may have somemore information about their system indicating that it is achievable.
If dissipation is increased to 5 W, the temperature at the mounting basereaches 155 °C which is probably not allowed. The alternative would be to
improve the thermal resistance to <4 K/W, which is extremely challenging.
An indication as to what can be achieved, is given in AN90003 (chapter 9 in thisbook) - LFPAK MOSFET thermal design guide.
The junction temperature of the MOSFET has not yet been mentioned. It isbecause it is only a few degrees higher than the mounting base. For example,
consider an application for an LFPAK56 device, such as BUK7Y7R6-40E. The
thermal resistance is 1.58 K/W. So for 2 W, the Tj would be 128.2 °C. For 5 W, it
would be 133 °C (assuming Tmb can be held to 125 °C). Both of these values arewell below the Tj(max) of the MOSFET which is 175 °C. So in summary, thelimiting factor of what can be done with dissipation is the PCB and its thermal
path to ambient, not the MOSFET.

372

8.4 Q: What is the position of Nexperia on
using Pb free solder for internal soldering(die attach, clip attach)?
A: Nexperia is a member of the DA5 working group. It is a project consortium
comprising Nexperia, Bosch, Infineon, and ST. The goal is to find new solder
materials or alternative die attach methods which do not use lead. The European
directive 2011/65/Eu exemption (RoHS), allows the use of lead in high melting
point solders until 2016. So far, no reliable and cost effective alternative process
has been developed, especially where the requirements of AEC-Q101 are considered. An extension to the expiry date of this exemption was applied for in January
2015 by representatives of the electronics industry, including the DA5 working
group.
The End of Life Vehicle (ELV) Directive (2000/53/EG) also applies. A similar
extension to the Pb free exemption was applied for by the DA5 group in November
2013. It is expected that if approved, this directive allows the use of Pb based
solders until 2018 at the earliest. It has been requested that the EU aligns the Pb
free exemption between the ELV directive and the RoHS directive.

9 SPICE models
9.1 Q: Is there is a large difference between
the data sheet and the SPICE model behavior
and in particular, the gate charge
characteristics?

13
Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

A: There is a strong similarity between the data sheet characteristics and the
Nexperia SPICE models at 25 °C. It is especially true for transfer curve, RDS(on), diodecharacteristic, and gate charge. The SPICE model also accounts for the packageparasitic resistances and inductances.
Additional information
In PWM circuits, the SPICE model gives quite a good similarity to the behaviorof the real device. The SPICE model can therefore be used to give a good
indication of the switching losses at turn-on and turn-off, as well as the
conduction losses.
The SPICE model is only correct at 25 °C, the RDS(on) versus temperature

373

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Nexperia | A Power Design Engineer's Guide

characteristic can be used to estimate conduction losses at higher
temperatures. Switching losses are known not to change significantly withtemperature.
The SPICE model also reflects a typical device according to the data sheet
characteristics.
The method of creating models has been continuously improved over time. The
latest model creation process used for Trench generation 6 devices and newer
technologies results in models which closely match measured device behavior.

9.2 Q: Why does the SPICE model not matchthe data sheet?
A: The SPICE models provided by Nexperia are generated from measurements
performed on a sample of devices. Several parameters such as transfer
characteristics, output characteristics and gate charge are used. Values for parasitic
package impedances and the data sheet maximum RDS(on) value are combined to
produce a model that emulates the behavior of the sample MOSFETs.
• It is important to note that the SPICE models generated by Nexperia:•
represent typical parts that can be found within the production 		
distribution.
•
are set close to the maximum RDS(on) of the part without adversely 		
affecting the other model parameters.
•
are only valid for Tj = 25 °C.
Customers wishing to do design validation using a SPICE model, are advised toproceed with caution given the information provided above. Nexperia encourages
designers to perform Monte Carlo simulations and use tolerance stacks in theirsimulation design. These factors permit part to part variation of their whole system
to be accounted for.
Nexperia can advise on what reasonable levels of tolerance on key parameters for
the MOSFET would be.

10 MOSFET silicon technology
10.1 Q: What is drift engineering?
A: Drift engineering is optimizing of the drift region between the bottom of the374trench and the epi/ substrate interface (light green area). The drift region supports
most of the drain-source voltage in the off state. The purpose of drift engineering
is to reduce the resistance of the drift region while maintaining the drain-source
breakdown voltage V(BR)DSS capability (see Figure 18).
SourceRpackage

SourceRmetal· Package type dependent· Leadframe/bond wires (clip)
resistance· Metal spreading resistance is layout andpackage dependent

Oxide
Gate

N
P

SwitchGate

Drift

RswitchSubstrate

Specific RDS(on)(mΩ mm2)

N
Drift Region
RdriftSubstrate

Rsubstrate

Drain

T1
Drain

T2

T3

T4

T6

Technology

aaa-016144

Figure 18 | Resistance in power MOSFETs

10.2 Q: What is obtained from reduced cell
pitch?
A: Reduced cell pitch generally results in lower resistance and higher capacitance.
The goal of each new generation of MOSFET technology is to reduce RDS(on) withouta large increase in capacitance that usually accompanies reduced cell pitch.
Reduced cell pitch also reduces SOA capability (linear mode operation) but
improves avalanche capability.

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

10.3 Q: What is obtained from a shorter
channel?
A: Shorter channel gives a lower RDS(on) and a lower CGS capacitance simultaneously.
It has higher leakage current and the transfer curve (ID versus VGS characteristic)
becomes more dependent on VDS. It is also observed in the output characteristics.

375

Power and small-signal MOSFET frequently asked questions and answers

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10.4 Q: What is obtained from thick bottomoxide?
A: Thick bottom oxide refers to gate oxide at the bottom of the trench (see Figure16). It is made thicker than the gate oxide at the side of the trench. It acts as athicker dielectric between the gate and the drain resulting in a much lower CGD
value.

11 Supply and availability
11.1 Q: What statements can be made concerning the long-term availability of previous
generations of TrenchMOS parts?
A: Nexperia continues to supply older products where the volumes of manufacture
are economically viable. The sales price margin is commercially viable and there areno manufacturing reasons which prevent manufacture.
A Discontinuation of Delivery (DoD) document notifies key customers (including
distributors), when a part is planned to be withdrawn. It allows customers to makearrangements to buy sufficient products for future requirements and if necessary
qualify alternative products.

12 EMC and ESD
12.1 Q: I have EMC issues, what can I do?
A: We have a detailed appliaction note on this subject, AN90011 (chapter 7 in thisbook) - Half-bridge MOSFET switching and its impact on EMC , please refer to thisfor any EMC related concerns.

12.2 Q: What parameters affect the ESD
tolerance and how much does it vary for aparticular device?
A: The key parameters are the gate oxide breakdown voltage and the gate input376

capacitance (Ciss). JESD22-A114 specifies the ESD Human Body Model test
arrangement and results assessment criteria.
Additional information
The main ESD failure mechanism for any MOSFET is due to breakdown of thegate oxide. The point where the oxide is thinnest will be where failure is most
likely to occur.
For Nexperia MOSFETs, factors which affect the thickness are:••Gate rating: usually standard level or logic level (note that other gate
oxide ratings are available from Nexperia)
Technology used: gate oxide thickness will vary according to the 		technology used. Check with Nexperia for details of the oxide thickness
or rating if this is critical information

The gate oxide is most sensitive between the gate and source. The gate to drain
path (typically the gate oxide at the bottom of the trench and semiconductor
junction in series) offers a higher capability.
There are 6 combinations of applying an ESD pulse to a MOSFET:

Table 3: ESD test pulses pin combinations
Pin ESD pulse applied toPin(s) grounded

Pin floating

G

S

D

G

S, D

-

D

G

S

D

G, S

-

S

D

G

S

D, G

-

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

The ESD pulse can be positive or negative for each condition. The simplified
capacitance model for the MOSFET shown in Figure 19 below is key tounderstanding how the gate oxide can be damaged by extreme voltage

CGD
NMOS
CDS
CGS

aaa-022333

Figure 19 | MOSFET schematic representing the internal capacitances377

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

The worst-case condition is always by applying the pulse to the gate pin with thesource pin grounded and the drain pin floating. This has been verified byexperiment. In the case of the pulse being applied between gate and source withthe drain floating, the capacitance of the device is minimised, see Figure 20 below.
Grounding the drain, for example, would increase the gate – source capacitance, so
it becomes more difficult to damage the oxide.
Note that the body diode acts to protect the drain pin of the device: if drain goes
negative due to ESD pulse on the drain, then diode conducts. The body diode iscapable of handling high current, much higher than due to ESD pulses so is notdamaged by ESD. If the drain goes positive, then the diode will avalanche and easily
absorb the ESD energy (like a TVS).
Drain floating

CGD
NMOS
CDS

Gate
ESD pulse

CGS

Sourceaaa-022333

Figure 20 | MOSFET schematic representing the internal capacitances

12.3 Q: How can gate-source ESD rating be
estimated?
A: This formula estimates the ESD capability:
Vesd (HBM) = 16 × VGS(max) × Ciss (nF)
Additional information
The simplest approximation for ESD is to assume that the device fails when Ciss ischarged to the breakdown voltage of the gate oxide (VBR(ox)). As a conservative
estimate VBR(ox) is approximately 1.6 times the VGS rating as per the limiting values
section in the data sheet.
Figure 21 shows the ESD test setup, once ESD test has completed then the voltage
across both capacitors is the same and total charge is conserved. At the last
surviving voltage, then voltage across the capacitors is as follows:
Vesd × Cesd = VBR(ox) × Cesd + VBR(ox) × Ciss
Vesd = VBR(ox) × (Ciss/Cesd + 1)

378

Vesd = 10 × VBR(ox) × Ciss
Vesd = 16 × VGS(max) × Ciss
RC

RD

M

charge current
limit resistor

VESD HIGH VOLTAGE
DC SOURCE

discharge
resistance

Ciss
CESD
100 pF

DeviceUnder
Teststorage
capacitor
aaa-031668

Figure 21 | Test circuit for ESD testing (human body model)

12.4 Q: Does relationship between Ciss and ESD
rating change by trench generation?
Yes. As per Section 12.3 the ESD rating relies upon Ciss and gate oxide breakdown
voltage. As Nexperia improves technology and the levels of quality and reliability
also improve new generations tend to have stronger gate oxides. However as we
improve our switching figure of merit (QG × RDS(on)), now for the same RDS(on) new
technologies will have lower Ciss and therefore lower ESD rating.

12.5 Q: Why are Nexperia power MOSFETs not
ESD protected on the chip by gate-source
protection networks?

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

In order to effectively screen MOSFETs with weak gate oxide and achieve <1 ppm
quality levels, Nexperia uses special test techniques which involve accurately
measuring the gate-source leakage behavior. Adding ESD protection networks
means that it becomes very difficult to measure the gate-source leakage
characteristics of the gate oxide because the ESD protection network will have asignificantly higher leakage current. This means we cannot screen out weakeroxides and will result in a higher field failure rates. Furthermore, adding protection
networks results in higher production costs. ESD protection networks are therefore
only used where necessary.
Generally, for larger MOSFETs with good gate oxide quality and relatively high Ciss

379

Power and small-signal MOSFET frequently asked questions and answers

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Nexperia | A Power Design Engineer's Guide

there is no need for ESD protection, as long as these are being mounted onto a PCB
in a controlled ESD environment. For special applications where the MOSFET would
be subjected directly to ESD in a finished product such as a lithium ion battery
module or a power or signal port then on-chip ESD protection may be required tomeet IEC 61000-4-2 or other ESD test specifications. Some very small MOSFETs
from Nexperia may require on chip ESD protection networks in order to allow
handling (such as NX3008NBKW), even in well controlled manufacturing
environments.

13 Leakage, breakdown and MOSFET
characteristics
13.1 Q: How does drain current (IDSS) vary withrespect to temperature?
A: The fundamental relationship between drain leakage current and temperature isexponential in form. The data sheet gives maximum values of IDSS at Tj = 25 °C and
175 °C. This example is specific to Nexperia Trench generation 2 technology but thesame principles can be applied for other Nexperia technology. An exponential fit tothese points provides the plot of Figure 22. It is also in line with some testing whichis performed during the development of new MOSFET technologies.
Figure 23 is the same curve, plotted with a log scale for IDSS to ease reading thevalue of 20 μA at 50 °C. These values are for a VDS at the rated voltage. Reducing
voltage reduces leakage current. Note that Figure 22 and Figure 23 are not in thedevice data sheets.
103

600

lDSS
(µA)

IDSS
(µA)
400

102

200

10

0

0

50

100

150

Tj (°C)

200

1

0

50

150

Tj (°C)200

aaa-013068

aaa-013067

380

100

Figure 22 | IDSS as a function of temperature,

Figure 23 | IDSS as a function of temperature,linear axes				log - linear axes

13.2 Q: What is the relationship between
breakdown voltage V(BR)DSS at ID = 250 μA) anddrain leakage current (IDSS)? Both state thesame VDSS value but the drain current isdifferent.
A: Although these two parameters reference the voltage rating of the part, theylook at different characteristics of the product. Drain leakage current (IDSS) is thecurrent which flows when VDS equal to the rated voltage is applied. The test checks
that the current is below the limit.
The breakdown voltage of a device V(BR)DSS is the VDS required to cause a draincurrent of 250 μA to flow. In practice it is slightly higher than the rated voltage ofthe device and the actual voltage varies for the same nominal type due tomanufacturing variations. The minimum V(BR)DSS stated in the data sheet is the
rated voltage. Breakdown voltage looks at the characteristic of the part when it isin avalanche. The mechanisms causing leakage current and avalanche current are
different.

13.3 Q: Is the standard level gate device
BUK7Y28-75B guaranteed to work with a 7 V
gate drive at -40 °C for 25 A?
A: Nexperia has a high degree of confidence that this scenario would be OK even in
the worst case. However, it cannot be 100 % guaranteed by a production test at
25 °C.
Referring to Figure 24, the typical gate threshold voltage VGS(th) is 3 V. It rises toapproximately 3.5 V at -55 °C (i.e. a rise of approximately 0.5 V). The highest VGS(th)rises from approximately 4 V to 4.5 V (again, approximately 0.5 V). So in the worst
case at -55 °C, the threshold voltage shifts by 1.5 V from the typical 25 °C value.

13
Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

Looking at Figure 25, it would shift the gate drive curve from 7.0 V to 5.5 V for aworst case device (1.5 V shift).

381

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

5
VGS(th)
(V)
4

max

3

typ

2

min

1

0
-60

0

60

120

Tj (°C)180

aaa-013069

Figure 24 | Gate threshold voltage VGS(th) as a function of temperature
100

120
RDS(on)
(mΩ)
100

10

ID
(A)

20

80

15

8

7

4.5

5

5.5

6

7

80
60

8

6
60

40

5
20

0

10
15

5.5

40

VGS (V) = 20 V

VGS (V) = 4.5

02

Tj = 25 °C

4

6

8

VDS (V)20

10

aaa-013070

0

0

20

Tj = 25 °C

40

60

80

ID (A)

100

aaa-013071

Figure 25 | Drain current as a function of

Figure 26 | Drain-source on-state resistancedrain-source voltage: typical values

as a function of drain current: typical values

The 5.5 V drive curve allows more than 25 A and is still in the linear RDS(on) region of
the output characteristic.
The RDS(on) values for the 5.5 V characteristic at 25 A in Figure 26 is pessimistic for
operation at -40 °C. It uses the 25 °C RDS(on) curves, and the carrier mobility increases with lower temperatures making the RDS(on) better. The threshold voltage
increase has already been accounted for.
It explains why a 7 V gate drive at -40 °C would be OK for this particular standard
level gate device (BUK7Y28-75B). The same principles can be applied to other
Nexperia devices. However, the customer must judge whether there is adequate
margin in the design, as the result may be slightly different from what is observed.

382

13.4 Q: What is the lowest voltage V(BR)DSS to
be expected at -40 °C for a 40 V device using
Trench generation 6?
A: The following principle could be applied to any Nexperia MOSFET technology at
any breakdown voltage rating. In the data sheet, the values for minimum drainsource breakdown voltages are specified at -55 °C and 25 °C. The correlation
between V(BR)DSS and temperature is approximately linear over this range.
Therefore, a straight line can be plotted at Temperature (-55 °C and 25 °C) versus
V(BR)DSS (at -55 °C and 25 °C).
For example: a 40 V Trench generation 6 part, has a V(BR)DSS at -55 °C of 36 V and 40
V at 25 °C. Using linear interpolation, gives a V(BR)DSS of 36.75 V at -40 °C.

13.5 Q: What factors affect the value of draincurrent according to the transfer characteristic
graph for BUK9275-55A, especially over the
VGS range of 2.2 V to 3.0 V?
A: The answer to this question is not simple - there are several factors which would
affect the ID value.
1. The graph depicted in Figure 27 is typical. The BUK9275-55A MOSFET has a
distribution of parameter values within the production tolerance limits. The
graph is only intended to illustrate how ID, VDS and VGS are related when the
MOSFET is operating for a particular VDS condition. VGS(th) has a significant
influence on the characteristic of a particular device. Temperature is also a major
factor. The limiting values and characteristics listed in the data sheet should be
used for circuit design.
2. Junction temperature Tj strongly affects the ID / VGS characteristic. The graph in
Figure 27 is for Tj = 25 °C and Tj = 175 °C. The same graph for the same part with
Tj = -55 °C, would be very different. The mode of operation preferred by thecustomer is with VGS in the range 2.2 V to 3 V in the saturation region before fullenhancement. In this mode, the MOSFET power dissipation is likely to be 		
significant. As a result, the junction temperature may be high. Figure 22 		
demonstrates how ID changes with Tj for a given VGS value.
3. A dashed vertical red line on the graph is shown at VGS = 2.2 V. If Tj increases
from 25 °C to 175 °C, ID approximately doubles (from approximately 1 A to 2 A).
However, at VGS = 2.8 V (dashed vertical green line), the same junction 		
temperature change has no effect on ID. At VGS > 2.8 V, an increase in Tj results
in a decrease in ID.

13Power and small-signal MOSFET frequently asked questions and answers

MOSFET & GaN FET Application Handbook

383

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

25
ID
(A)
20

15

10

5

0

Tj = 175 °C

01

Tj = 25 °C

2
3
VDS = 25 V

4

VGS (V)5

aaa-015948

Figure 27 | Transfer characteristics: drain current as a function of gate-source voltage - typical values

It is clear that the relationship between the MOSFET parameters is complex andtheir relationship with the thermal environment is also complex.

13.6 Q: Can Nexperia provide CGD, CGS and CDS
numerical values for Tj = -55 °C and Tj = +175 °C(at VGS = 0 V, VDS = 16 V)? If it is impossible totest, a theoretical one is also acceptable. A
graph is provided in Figure 23, but it is for Tj =+25 °C.
A: Unfortunately, Nexperia cannot supply values for these capacitances at theextremes of the MOSFET operating temperature range requested. It is due to thelimitations of our parametric test equipment. However, we can comment on how
these capacitances vary with temperature and the MOSFET terminal voltages.
Ciss is the input capacitance formed by the parallel combination of CGS and CGD, and
CGS dominates. CGS is formed across the gate oxide so it does not vary significantlywith temperature or the MOSFET terminal voltages. As CGS depends on gate oxide
thickness and other defined die feature dimensions, it should not vary much
between samples.
Crss is the reverse transfer capacitance which is essentially the gate-drain
capacitance (CGD). It is formed across the MOSFET body diode depletion layer. This
layer becomes thicker, as the reverse voltage (VDS) across it increases. Crss increases
as VDS decreases. Crss has a greater variability than Ciss because it depends on the
body diode depletion layer.

384

Coss is the output capacitance formed by the parallel combination of CDS and CGD.
The drain- source capacitance (CDS) also dominates this capacitance. It varies with
VDS in a similar way to Crss varying with VDS and it has similar variability to Crss for thesame reasons.
These relationships are illustrated on the data sheet graph depicted by Figure 28. It
has been observed that switching losses only slightly increase at Tj(max), in the orderof 10 %, since the capacitances only marginally change. Other factors can influenceswitching behavior, especially where the gate driver current capability changes
significantly with temperature. The depletion layer thickness varies in proportion to
the square root of the absolute temperature in K and it affects Crss and Coss.
104

C
(pF)

Ciss

103
Coss
Crss

102
10-1

1

10

VDS (V)102

aaa-013072

Figure 28 | Input, output and reverse transfer capacitances as a function of drain-source voltage:
typical values, VDS = 0 V

13.7 Q: Can Nexperia provide the minimum VGS
threshold values for Tj from -55 °C to +175 °C?
A graph is already in the data sheet. However,the numerical data of minimum values in therange of -55°C to +175 °C are required for astandard level gate threshold.

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application Handbook

A: Worst case values of minimum and maximum VGS(th) should be used for design385

Power and small-signal MOSFET frequently asked questions and answers

13

Nexperia | A Power Design Engineer's Guide

purposes. They are given in the data sheet Characteristics (see Table 4).
Additional information

Table 4: Characteristics
Voltages are referenced to GND (ground = 0 V).
Symbol Parameter

VGS(th)gate-source thresholdvoltage

Conditions

Min

Typ

Max Unit

ID = 1 mA; VDS = VGS;
Tj = 25 °C

2.4

3

4

V

ID = 1 mA; VDS = VGS;
Tj = -55 °C

-

-

4.5

V

ID = 1 mA; VDS = VGS;
Tj = 175 °C

1

-

-

V

These values are guaranteed worst case values. In this case, the 175 °C minimum
VGS(th) is not less than 1 V. The -55 °C VGS(th) is not greater than 4.5 V.

13.8. Q: Can Nexperia provide the inherent RG
component value with Tj from -55 °C to+175 °C?
A: The measured RG value is in the range of 1 Ω to 3 Ω and it does not varysignificantly with temperature. In our general MOSFET characterization, it ispresently not possible to test RG over the temperature range.
Additional information
In a circuit such as the 3-phase motor drive circuit, switching speed is not usually
critically important. The PWM frequency is usually moderate (<50 kHz).
However, to mitigate emissions due to high dV/dt, the circuit designer often
deliberately slows the switching of the MOSFET.
A low value (10 Ω) fixed resistor connected between the gate driver output andthe MOSFET gate helps to stabilize the gate driver voltage and damp out any
voltage transients or oscillations.
Often, even higher external gate driver resistor values are chosen to slow down
the gate driver and reduce the EMI effects of the MOSFET switching.

386

13.9 Q: How is the maximum permissible draincurrent estimated, at the point of VDS = 0.1 V,from the SOA curve of the BUK9K12-60E
shown in Figure 24? Some manufacturers
describe down to the VDS value in their SOA
curves.
103
ID
(A)
102

Limit RDS(on) = VDS / ID
tp = 10 µs
100 µs

10
DC
1

1 ms
10 ms
100 ms

10-1

10-1

1

102

10

VDS (V)103

aaa-022335

Figure 29 | Safe operating area: continuous and peak drain currents as a function of drain-source
voltage

A: The minimum current that is expected at a VDS of 0.1 V can be calculated fromthe maximum (175 °C) RDS(on) value (26 mΩ):

Table 5: RDS(on) values
Symbol Parameter

RDS(on)drain-source on-state
resistanceConditions
VGS = 5 V; ID = 15 A; Tj =25 °C
VGS = 5 V; ID = 15 A; Tj =175 °CMin

Typ

Max Unit-

9.5

11.5 mΩ-21.5 26

Power and small-signal MOSFET frequently asked questions and answers

13

MOSFET & GaN FET Application HandbookmΩ

The drain current that flows with these conditions is 0.1/0.026 = 3.846 A. The
maximum die temperature is the critical factor. Do not allow it to exceed 175 °C.
However, if the RDS(on) is not at the top limit of the value range or the die temperature is lower, it is lower. As a result, the corresponding drain current is proportionately higher.
From Table 5 the maximum RDS(on) is 11.5 mΩ at Tmb = 25 °C. The maximum die

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temperature is likely to be higher than 25 °C in most applications.
If the mounting base temperature is maintained at 100 °C or less, the (fully ON)
MOSFET can safely carry a continuous current up to 35 A, (see Table 6).

Table 6: ID values
Symbol ParameterIDdrain current

Conditions

Min

Typ

Max Unit

Tmb = 25 °C; VGS = 5 V

-

-

35

A

Tmb = 100 °C; VGS = 5 V

-

-

35

A

The (fully ON) MOSFET can also sustain a current pulse of 204 A for a period up to
10 μs, (see Table 7).

Table 7: IDM values
Symbol Parameter
IDM

peak drain current

Conditions

Min

Typ

Max Unit

Tmb = 25 °C; tp ≤ 10 μs

-

-

204

A

13.10 Q: Is the BUK7K52-60E drain current
specified individually for each (FET1 and FET2)
or as a total of 15.4 A?
Table 8: Drain specifications
Symbol Parameter

Conditions

Min

Typ

Max Unit

VDS

drain-source voltage

Tj ≥ 25 °C; Tj ≤ 175 °C

-

-

60

V

ID

drain current

VGS = 10 V; Tmb = 25 °C--

15.4 A

Ptot

total power dissipation

Tmb = 25 °C

-

-

32

W

A: The ratings given on the data sheet are for each individual MOSFET in thisdevice, (see Table 8).
Although there are two MOSFETs housed within the package, they are fully
electrically isolated from each other.
However, as the MOSFETs share a common package, there is a small amount of388

thermal coupling between the two MOSFET dies through the plastic package
material. The heat generated by the power dissipated in one MOSFET increases thetemperature of the other, even though the other may not be dissipating power. In
an application, there is also an external thermal coupling path via the PCB to which
the device is mounted. In practice, it is the main thermal coupling mechanism
between the two dies.
To guarantee long-term reliability, it is very important that the junction temperature of either of the dies is never allowed to exceed 175 °C.
The individual MOSFET mounting bases are the main exit routes for heat generated
in the dies. In practice, the mounting bases are soldered to copper pads on a
Printed-Circuit Board (PCB). They provide the electrical connections to the MOSFET
drains and heat sinking. Both MOSFETs in the package should operate at their rated
power/current when their mounting bases are maintained at 25 °C. However, it isvery difficult to achieve in practice and de-rating must be done in most cases.
Example:Using a BUK7K52-60E for two MOSFETs in a half-bridge, driving an inductive load.
Assumptions:
MOSFETs conduct alternately, never together.
Switching losses are neglected.
PCB material is standard FR4.
Standard FR4 PCB material has a maximum operating temperature of 125 °C,
although special higher temperature materials are available.

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MOSFET & GaN FET Application Handbook

The maximum temperature of the MOSFET mounting bases is limited to 125 °C byappropriate PCB cooling. The maximum de-rated power that can be dissipated in
one MOSFET is:
32/3 = 10.67 W.
Hence, the temperature rise of the die above the temperature of its mounting baseis: 10.67 (W) × 4.68 (K/W) = 49.9 K.
Therefore, the maximum die temperature is 125 + 49.9 = 174.9 °C, which is
marginally safe.
Worst case RDS(on) at Tj = 175 °C = 101 mΩ.

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Maximum allowed current = √(P/RDS(on)) = 10.28 A.
Note: This calculation is a theoretical example. The PCB cooling required tomaintain a 125 °C mounting base temperature with 10.67 W power dissipation
would be very difficult to achieve in practice.

13.11 Q: How does V plateau shift withtemperature and process variation?
Data from a T9 MOSFET family device BUK7J1R4-40H is considered but the
principle can be applied to T6 devices also. The plateau voltage in the gate charge
characteristic is the horizontal portion of the graph (Figure 30) and is related to thetransfer characteristic (Figure 31).
VGS
(V)

360
ID
(A)
300

10

8

240
6
VDS = 14 V

180

32 V

4
120
2

0

25°C

60

0

10

20

30

40

Tj = 25 °C; ID = 25 A

50

60
70
QG (nC)80

aaa-026115

0

175°C
Tj = -55°C
0

1

2

3

4

VDS = 8 V

56
VGS (V)7

aaa-026113

Figure 30 | Gate-source voltage as a function of

Figure 31| Transfer characteristics; drain

gate charge; typical values

current as a function of gate-source					voltage; typical values

The plateau voltage is around 4.25 V typical for a current of 25 A. This corresponds
to the value in the transfer curve, also for a typical device. So at -55 °C then theplateau voltage will be 4.35 V and at 175 °C it will be 3.9 V for a typical device.
Note: transconductance reduces with increasing temperature, so the device
will switch a little slower and switching losses will increase.
When considering a “worst case” device then the spread in gate threshold VGS(th)needs to be considered. It is assumed that the gain (transconductance) of thedevice is not affected by the same process related reasons which affect VGS(th) . The
transfer curve for a typical device would be shifted along the VGS axis according to390

the delta in the VGS(th) (Figure 32).
The plateau voltage at the 25 A test condition would be 3.65 V for minimum VGS(th)and 4.85 V maximum VGS(th) .

Figure 32 | Transfer characteristic at 25 °C

13.12 Q: What is the current which each device
can conduct and how does this relate to thedevice power rating for the SOT1205 device?
A. Consider a specific example such as BUK9K52-60E. The data sheet shows thefollowing capability:

Table 9: Quick reference data; BUK9K52-60E
Symbol Parameter

Conditions

Min

Typ

Max Unit

VDS

drain-source voltage

Tj ≥ 25 °C; Tj ≤ 175 °C

-

-

60

V

ID

drain current

VGS = 5 V; Tmb = 25 °C

-

-

16

A

Ptot

total power dissipation

Tmb = 25 °C

-

-

32

W

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The key point is the Ptot of 32 W. This is per die at data sheet conditions whichassume that the mounting base is maintained at 25 °C. The maximum DC current
allowed in each device would be 16.04 A, based on RDS(on) of 124.3 mΩ (VGS= 5 V) at
175 °C.

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IDC = (PIR)
IDC = (32/0.1243) = 16.04 A
If both devices in the package are considered then the total power dissipation
when both mounting bases are maintained at 25 °C is 32 W x 2 = 64 W. This only
applies when the mounting bases of the devices are maintained at 25 °C (using an
infinite heatsink). The power capability will decrease as the mounting basetemperatures increase such that Tj does not exceed 175 °C. Consequently thecurrent will decrease as shown in the graph Figure 33 of ID vs Tmb, if the mounting
base is maintained at a different temperature such as 125 °C, the current rating
would be 9.26 A.

ID
(A)

20

16

12

8

4

0

0

30

60

90

120

150
Tmb (°C)

180

VGS ≥ 5 V

003aaj559

Figure 33 | Continuous drain current as a function of mounting base temperature

In practice it is not a realistic condition to mount the device on an infinite heatsink,the device might be mounted on a PCB with Rth(mb-amb) = 50 °C / W. In which case
how can the steady state current capability be calculated?
The diagram in Figure 34 below shows a simplified thermal model from thejunction of each die to ambient. It is assumed that there is good coupling on thesubstrate such as a PCB between the devices (usually the case but not always) andthat there is minimal coupling between the devices within the package (this is>100 °C / W).

392

Tj2

Tj1
Rth j1 to j2
100 K/W

Rth j1 to mb1
4.7 K/W

Rth j2 to mb2
4.7 K/W
Rth mb1 to mb2
5 K/W

Tmb1

Tmb2

Rth mb2 to amb

Rth mb1 to amb
100 K/W

100 K/W

Tamb

Tamb

Figure 34 | Dual MOSFET thermal configuration

If the ambient temperature is 25 °C then the maximum total package power
dissipation would be 2.87 W assuming the power is shared equally between thetwo dies.
P x 100 + P x 4.7 = T Δ = 150 ºC
P = 1.433 W per die

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This corresponds to a maximum current of 3.40 A in each device simultaneously.
IDC = (PIR)
IDC(1.433/0. 1243) = 3.40 A

This will result in a mounting base temperature of 168.2 °C. This is above the typicaltransition temperature for the PCB of 125 °C. The simplified thermal resistancemodel can be used to calculate the power based on the actual conditions andrequirements. For highly accurate calculations, computational fluid dynamics
simulations or calibrated measurements should be conducted.

393

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14 MOSFET reliability
14.1 Q: How is the FIT-rate calculated?
A: FIT (Failure In Time) is commonly used to express component reliability. It isdefined as the number of failures occurring in 1 × 109 hours (1 billion hours).
At any elapsed time (t), the reliability (R) of a group of operating semiconductors is:
R(t) = (no - nf)/no
Where:
no is the original sample size and nf is the number of failures after time t.
Over the standard time of 109 hours, it approximates to F = (1/no)*(nf/t)*109.
Accelerated testing
A major factor in determining the reliability of a semiconductor is the total
stress applied by the application. Operating temperature results from ambient
temperature and the heat due to power dissipation. It is the most important
applied operating stress where a product is otherwise generally operated
within its ratings.
The Arrhenius equation is used to model the effect of temperature on
component failure rate:
Acceleration factor = e(Ea/k)(1/T1 - 1/T2)			

[7]

Where:
Ea = activation energy (eV)k = Boltzmann constant (8.60 × 10-5 eV / K)
T1 = operating temperature (°C)
T2 = reliability test temperature (K referenced to absolute zero)
Accelerated testing makes components perform at high levels of (thermal) stress.
The results are then extrapolated to convert short life under severe conditions into394

the expected life under normal conditions.
Under accelerated life test conditions:
F = (nf/n0 × A × t) × 109 FITs					

[8]Time t is now equal to A × t, where A is the acceleration factor.
Based on the life-test results for an example part, the FIT data provided in Table 10
has been calculated. Note that an adjustment is made to the number of failures
based on Poissons probability distribution. It is used to indicate the number of
failures expected in a larger sample depending on the confidence level. It is
described in JEDEC JEP122F Section 5.18.1.4 and in numerous other references.

Table 10: Calculated FIT data

Test nameHigh Temperature Reverse Bias (HTRB) +High Temperature Gate Bias (HTGB) +High Temperature Storage Bias (HTSL)Test temperature

175

Number of device hours

35,051,000°C

Number of observed failures 0
Confidence level

90%

Activation energy

0.7

eV

Failure rate

Unit

Failure rate at

125

°C

at 90 % confidence =

6.725

FIT

Failure rate at85°C

at 90 % confidence =

0.688

FIT

Failure rate at55°C

at 90 % confidence =

0.086

FIT

Failure rate at25°C

at 90 % confidence =0.0007FIT

MTBFPower and small-signal MOSFET frequently asked questions and answers

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MOSFET & GaN FET Application Handbook

Unit

Mean time before failure at

125

°C

at 90 % confidence =1.49 × 108

hour

Mean time before failure at85°C

at 90 % confidence =1.45 × 109

hour

Mean time before failure at55°C

at 90 % confidence =1.16 × 1010

hour

Mean time before failure at25°C

at 90 % confidence =1.40 × 1011

hour

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15 References
1. AN11158 — Understanding power MOSFET data sheet parameters
2. AN10273 — Power MOSFET single-shot and repetitive avalanche ruggedness
rating3. AN90003 — LFPAK MOSFET thermal design guide
4. AN11160 — Designing RC snubbers
5. AN90001 — Designing in MOSFETs for safe and reliable gate-drive operation

16 Tables
Table 1.
Table 2.

Turn on/off delay, rise time and fall time requirements..........................369
Turn on/off delays, rise time and fall timeBUK9Y29-40E specifications.........................................................................370
Table 3. ESD test pulses pin combinations................................................................377
Table 4. Characteristics.................................................................................................386
Table 5. RDS(on) values....................................................................................................387
Table 6. ID values............................................................................................................388
Table 7. IDM values..........................................................................................................388
Table 8. Drain specifications........................................................................................388
Table 9. Quick reference data; BUK9K52-60E..........................................................391
Table 10. Calculated FIT data.........................................................................................395

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17 Figures
Figure 1. Comparison of transient thermal impedance............................................341
Figure 2. MOSFET thermal resistance junction to ambient.....................................344
Figure 3. LFPAK88 (BUK7S1R0-40H) and LFPAK56E(BUK7J1R0-40H)
transient thermal impedance comparison.................................................347
Figure 4. Diode characteristics......................................................................................348
Figure 5. Example: BUK7Y12-55B SOA curve showing derating for 100 °C..........350
Figure 6. Normalized total power dissipation as a function of mounting
base temperature...........................................................................................351
Figure 7. Safe operating area: continuous and peak drain currents as afunction of drain-source voltage.................................................................352
Figure 8. Simplified planar and trench technology...................................................354
Figure 9. Avalanche safe operating area.....................................................................361
Figure 10. Power MOSFET showing two possible
mechanisms for dV/dt induced turn-on......................................................365
Figure 11. Physical origin of the parasitic BJT
components that may cause dV/dt induced turn-on................................365
Figure 12. Trench generation 6 structure.....................................................................366
Figure 13. Blue VGS trace shows gate bounce and the red current
waveform spiking when gate bounce occurs............................................366
Figure 14. MOSFET Schematic representing the internal capacitances..................367
Figure 15. Simplified schematic of a MOSFET half- bridge. The high-side .....................
and low-side MOSFETs are switched antiphase to each other...............367
Figure 16. Waveforms showing the full detail of the gate bounce
phenomenon...................................................................................................368
Figure 17. Effect of RG on VGS, and shoot-through current........................................369
Figure 18. Resistance in power MOSFETs.....................................................................375
Figure 19. MOSFET schematic representing the internal capacitances...................377
Figure 20. MOSFET schematic representing the internal capacitances...................378
Figure 21. Test circuit for ESD testing (human body model).....................................379
Figure 22. IDSS as a function of temperature, linear axes............................................380
Figure 23. IDSS as a function of temperature, log - linear axes...................................380
Figure 24. Gate threshold voltage VGS(th) as a function of temperature..................382
Figure 25. Drain current as a function of drain-source voltage: typical values.......382
Figure 26. Drain-source on-state resistance as a function of drain current:
typical values...................................................................................................382
Figure 27. Transfer characteristics: drain current as a function ofgate-source voltage - typical values............................................................384
Figure 28. Input, output and reverse transfer capacitances as a function ofdrain-source voltage: typical values, VDS = 0 V...........................................385
Figure 29. Safe operating area: continuous and peak drain currents as afunction of drain-source voltage.................................................................387
Figure 30. Gate-source voltage as a function of gate charge; typical values..........390

Figure 31. Transfer characteristics; drain current as a function ofgate-source voltage; typical values..............................................................390
Figure 32. Transfer characteristic at 25 °C....................................................................391
Figure 33. Continuous drain current as a function of mounting basetemperature....................................................................................................392
Figure 34. Dual MOSFET thermal configuration..........................................................393

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Chapter 14
Leakage ofsmall-signal
MOSFETs
Application Note: AN90009

Leakage of small-signal MOSFETs

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1 Introduction
Description of current leakage behavior for small-signal MOSFETs. Examples fortypical gate- source and drain-source leakage of MOSFETs is shown. Distinguishbetween ESD protected and unprotected devices.
Device current leakage is an important factor for the energy consumption of acircuit application. This is especially true for mobile electronic devices such as smart
phones, tablets, wearables or medical devices as the off-state current leakage may
influence battery life and therefore operation time. The scope of this applicationnote is the current leakage description of a small-signal trench MOSFET in different
pin configurations.

2 Leakage currents
The following chapters are related to the pin configurations drain-source (D-S) andgate-source (G-S) as illustrated in Figure 1 below.
D

D
G

VDS

VDS

G
±VGS

±VGS
S

S

a. ESD protected small signal MOSFET

b. Non protected small signal MOSFET

aaa-030427

aaa-030428

Figure 1 | Illustrated pin configuration for device leakage

For gate-source leakage it is important to distinguish between ESD protected
MOSFETs and non- protected ones. Due to the presence of additional ESD
protection circuit the VGS to IGS leakage characteristic of such a device is different
while the drain-source characteristic is independent from this constructionalfeature.

2.1 Drain-source leakage
For an enhancement mode MOSFET in the off-state (VGS = 0 V), the drain-source
leakage is given by a p-n junction diode in reverse direction. In the ideal case this isgiven by the rectifier equation:402

I = IS [ exp ( qV / kT ) - 1] ≈ IS

(V < 0, - V < < kT / q)

		

[1]

Where V is the applied voltage, q the electronic charge, k the Boltzmann constantand T the absolute temperature. V > 0 represents the diode in reverse direction
with a remaining IS, (the so called saturation current):

		

[2]

with the cross-section area A, the diffusion coefficients of holes and electrons Dp,
Dn, the donor and acceptor concentrations ND, NA, the intrinsic carrier
concentration ni and the carrier lifetime of holes and electrons τp.
It can be seen that the ideal description of the leakage is independent from theapplied voltage. The leakage increases with the cross section area A and therefore
the channel width which is proportional to the Xtal size if the same technology isused. The larger the Xtal the higher the leakage.

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MOSFET & GaN FET Application Handbook

In practice an additional current is added to the ideal saturation current which isincreased at higher reverse voltage applied. (Moll, 1958) Reasons for thiscomponent are:
• Surface leakage and inversion layers• Body defects• Generation and recombination
10-3
ID
(A)
10-4

VDS(max)

10-5
IDSS(max)

10-6
10-7

PMCM6501VPE
10-8
PMCM4401VPE
10-9
10-10

0

5

10

15

VDS (V)20

aaa-030429

Figure 2 | Drain-source ID leakage current as a function of drain-source voltage for PMCM4401VPE
and PMCM6501VPE at Tj = 25 °C

403

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The drain-source leakage as a function of drain-source voltage is given by Figure 2.
Both types have a specified maximum voltage of 12 V with a safety margin up to
the breakdown voltage of approximately 17 V when avalanche occurs. Specified
maximum value for IDSS leakage at maximum drain source voltage specified is 1 μA
however in reality for small-signal MOSFET this current is significantly below thisstandardized limit. Due to the larger chip size by a factor of 2.3 the PMCM6501VPE
has an increased leakage which is related to the higher channel width described inequation (2). Even for the larger chip the real leakage is more than one order ofmagnitude away from the specification limit.
The saturation current IS is temperature dependent. In case of silicon therecombination and generation of carriers is increased above room temperatureleading to increased leakage. This relation is given as example for PMCM4401VNE
in the figure below:
10-4
ID
(A)
10-5

10-4
ID
(A)
10-5
150 °C

10-6

10-6

10-7

10-7
70 °C

10-8
10-9
10-10

10-8
10-9

25 °C

0

5

10

15

20
25
VDS (V)a. As a function of drain-source voltage; VGS = 0 V
aaa-030430

10-10

0

25

50

75

100

125

150
175
Tj (°C)b. As a function of junction temperature;
VDS = 12 V, VGS = 0 V

aaa-030431

Figure 3 | Drain-source leakage current of PMCM4401VNE

2.2 Gate to Source leakage
An unprotected MOSFET has an electrically isolated gate with very small leakage
from gate to source while a protected MOSFET shows the typical leakage curve of abidirectional Zener diode. Figure 4 below shows the gate-source leakage current of
the ESD protected BSS138BK and the unprotected BSS138P. Both types are
specified with a maximum gate to source voltage of 20 V. Channel width and gate
oxide are equal.

404

10-4
IG
(A)
10-5

0.001
IG
(A)
0.0008

BSS138BK

10-6
BSS138BK

0.000610-7
10-8

0.0004

BSS138P

10-9

BSS138P

10-10

0.0002

10-11
0

0

10

20

30

10-12

40
50
VDS (V)

0

10

20

30

40
50
VGS (V)

aaa-030432

aaa-030433

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MOSFET & GaN FET Application Handbook

Figure 4 | Gate-source leakage current of ESD protected BSS138BK and non protected BSS138P

It can be seen that where minimizing gate-source leakage is essential an
unprotected MOSFET gives the best choice over a wide voltage range. The Zener
diode of a protected device is a trade-off between ESD protection and device
gate-source leakage. It is designed in a way that the clamping voltage is lower thanthe oxide breakdown voltage to protect the ESD most vulnerable part of the
MOSFET and higher than the specification to keep the leakage at a minimum over
specified range. Due to the poly-crystalline construction of the Zener diode theleakage below breakdown voltage is still higher compared to a single crystal diode.
The gate leakage temperature dependency for the Zener diode is similar to a silicon
p-n junction. The following figure shows an increase in leakage for higher
temperature. The leakage of an un-protected MOSFET remains almost constant in
the same temperature range.
10-5
IGS
(A)

150 °C

10-6

70 °C
25 °C

10-7

10-8

10-9

10-10

0

VGS(max)= 8 V

2

4

6

8

10

VGS (V)12

aaa-030434

Figure 5 | Gate-source leakage current of an ESD protected MOSFET at different junction
temperatures405

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3 Applications
Normally, when selecting a MOSFET for an application, the typical selection criteria
are things like VDS voltage, RDS(on), gate threshold voltage and so on. With theongoing trend towards battery driven and even energy harvesting based
applications, other performance criteria such as leakage start to be become more
important.
There are several use-case scenarios where ultra-low leakage of a MOSFET is a key
parameter and one of the first things a designer looks at when selecting thecomponent.

3.1 Load switch
When powering an application from a finite power source such as a battery,
designers often implement so called low power modes of operation. Most modern
microcontrollers feature several levels of low power operational modes.
Nevertheless, to get the lowest possible application standby power, the best way is
still to simply power everything down while leaving a small fraction of the circuit
powered up to handle detection for waking back up. The whole application
operates then in a PWM mode of operation. It is powered off most of the time,
while only waking up periodically.
The common method to implement this powering-up and down is to use a loadswitch that (dis)connects the circuit power from the battery. Many load-switch IC
solutions exist, often offering additional features such a current limit, in-rush limit
and more. Still, the simplest way to implement a load-switch function is to use a
MOSFET. Refer to Figure 6. Here we see a high-side switch implemented with aP-channel MOSFET.
Battery
R1P-channel MOSFET

Application
Power Plane

R2

CTRL

N-channel MOSFET

aaa-030435

Figure 6 | Load-switch using MOSFETs

406

One of the nice “side-effects” of using a MOSFET for this function is that theleakage current can be very low. Clearly when implementing a switch to turn somecircuit completely off, you would want that switch to be as high Ohmic as possible,
i.e. the leakage current between source and drain should approximate zero as
much as possible.
Per Figure 6 a separate N-channel MOSFET is often used to control the P-channel
MOSFET. A high level on the gate of the N-channel MOSFET will pull the gate of the
P-channel MOSFET low and turn it on (connecting battery to the application power
plane). Here we see another important leakage path, the one from the control
drive line via the gate of the N-channel to either Source or Drain. For optimal
application standby time an N-channel MOSFET with very low gate leakage
is required. Resistors R1 and R2 are used to further optimize leakage levels. Theirvalues are a trade-off between lowest leakage levels obtainable, VGS switch levels
reachable and switching speed of the load-switch circuit.

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MOSFET & GaN FET Application Handbook

3.2 General purpose logic
For those applications that require the absolute lowest leakage, it may actually bebetter to implement basic logic functionality (such as gates) using discrete
MOSFETs. While not very commonplace, this is sometimes required in applications
using energy harvesting as their only source of power.
The exact circuit implementation dictates what is required, but refer to Figure 7,
where a basic discretely build inverter is shown (simplified). Clearly, for lowest
possible system level leakage, gate-source and drain-source leakage are importanthere.

P-channel MOSFETIN

OUT
N-channel MOSFET

aaa-030436

Figure 7 | Inverter using MOSFETs (simplified)407

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4 Selecting a low leakage MOSFETThe data as shown in this note elaborates on the (physical) background of leakage
in MOSFETs. To help with selecting the right product for low leakage applications,
we can summarize this as follows:• As the drain-source voltage across the MOSFET gets closer to the maximum
specified VDS there is a risk of significantly increased channel leakage. Hence theneed to stay away from the maximum specified VDS• At higher temperatures, leakage goes up significantly and not the same for all
MOSFETs. Low threshold parts will see a bigger increase• With rising temperature, leakage increases. Leakage will increase faster in low
VGS(th) devices• Gate protection structures cause an increase to gate leakage.
Combining all of the above, we can argue that the highest chance for lowest
possible leakage is to select a device that:
1. Has a max VDS that is at least twice as high the highest VDS that will be seen in
the application
2. Has the highest acceptable VGS(th) to help minimize leakage increases at higher
temperature
3. Does not have protection on the gate. While not ideal for ESD sensitivity, an
unprotected gate will help keep gate leakage down, especially at elevatedtemperatures
4. Has the highest acceptable RDS(on). While not 100% accurate, if we assume thesame process for two different devices, the type with the lower RDS(on) has alarger die size, and hence will show a (minimally) higher leakage.
In all cases Nexperia can only guarantee the limit as per specification.

5 Nexperia low leakage MOSFET portfolio
The market shows a large and growing interest in MOSFETs with a low leakage
specification for IDSS and IGSS due to an increasing number of battery-driven
applications in the market such as wearables, Bluetooth trackers, wirelessmeasurement devices, computing accessories, e-metering and smoke detectors. In
applications that are required to have a long battery life time, although driven by asmall coin-cell, every additional leakage current has to be avoided.
Nexperia introduced a low leakage MOSFET portfolio depicted in Table 1 below.
The space saving leadless packages DFN1006-3, DFN1006B-3 and DFN1010B-6
have been chosen to support compact board designs. The height of DFN1006-3 is408

0.48 mm, DFN1006B-3 and DFN1010B-6 have a reduced height of 0.37 mm only. In
DFN1006 single FETs as 20 V P-channel and N-channel version are available. The
DFN1010 variants are dual-FETs containing a dual N-channel, a dual P-channel or acomplementary pair. Using the complementary PMCXB900UEL allows for a very
compact load switch, with the topology of a high-side switch shown in Figure 6. For
small battery-driven applications, the load currents are normally not very high sothat an RDS(on) of about 1 - 2 Ohms is fully sufficient.

Table 1: Low leakage current MOSFET portfolio

Type name

Polarity VDS VGS ID

RDS(on)@ VGS= 4.5 VRDS(on)@ VGS= 2.5 VRDS(on)@ VGS=1.8 V

(V)

(mΩ)

(mΩ)

(mΩ)

(V)

(A)

Package

PMZ600UNEL

N

20

8

0.6

470

620

845

DFN1006-3

PMZB600UNEL

N

20

8

0.6

470

620

845

DFN1006B-3

PMZ950UPEL

P

20

8

0.5

1020

1270

1700

DFN1006-3

PMZB950UPEL

P

20

8

0.5

1020

1270

1700

DFN1006B-6

20

8

0.6

470

620

845

DFN1006B-6

PMDXB600UNEL dual N
PMDXB950UPEL dual P
PMCXB900UEL

20

8

0.5

1020

1270

1700

DFN1006B-6

N

20

8

0.6

470

620

845

DFN1006B-6

P

20

8

0.5

1020

1270

1700

DFN1006B-6

Leakage of small-signal MOSFETs

14

MOSFET & GaN FET Application Handbook

The maximum leakage for the drain -source path is reduced from 1 µA down to
25 nA specified for VDS equal 5 V. The gate-source leakage limit is lowered from
10 µA to 50 nA for a gate voltage of 1.8 V.

Table 2: Comparison of maximum leakage current ratings between
standard and low leakage MOSFETs
Standard typeLow leakage type

IDSS(max)

1 μA

25 nA @ VDS = 5 V

IDSS(max)

10 μA

50 nA @ VDS = 1.8 V

6 References
Moll, J. L. (1958). The Evolution of the Theory for the Voltage-Current
Characteristic of P-N Junctions. Proceedings of the IRE (Volume:46 , Issue: 6 ),
1076 - 1082.

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7 TablesTable 1.
Table 2.

Low leakage current MOSFET portfolio.....................................................409
Comparison of maximum leakage current ratings between
standard and low leakage MOSFETs ..........................................................409

8 Figures
Figure 1. Illustrated pin configuration for device leakage.......................................402
Figure 2. Drain-source ID leakage current as a function of drain-source...............403
voltage for PMCM4401VPE and PMCM6501VPE at Tj = 25 °C...............404
Figure 3. Drain-source leakage current of PMCM4401VNE.....................................405
Figure 4. Gate-source leakage current of ESD protected BSS138BK
and non protected BSS138P........................................................................405
Figure 5. Gate-source leakage current of an ESD
protected MOSFET at different junction temperatures..........................405
Figure 6. Load-switch using MOSFETs.........................................................................406
Figure 7. Inverter using MOSFETs (simplified)...........................................................407

14
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Chapter 15
DC-to-DC conversion
with small-signal
MOSFETs
Application Note: AN11119

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1 Introduction
This chapter explores different methods of DC-to-DC conversion.
It contains several examples of DC-to-DC down-converters using small-signal
MOSFETs.
In modern electronic designs various supply voltages need to be generated. In mostcases power supply provides one or a few different DC voltages. These voltages
need to be converted to another voltage for several functional units in theapplication. Voltage conversion can work in both directions: it can be step up or
step down.
Using linear voltage regulators for voltage conversion was common, even if avoltage had to be reduced significantly or if the load currents were high. Because
linear voltage regulator works as a controlled series resistor, a lot of energy is
dissipated thermally.
Due to the environmental requirement to improve energy efficiency of electronicequipment, switch mode power supplies are replacing linear voltage regulators. In
switch mode power supplies, energy is stored in the magnetic field of inductors or
as a charge in capacitors. Ohmic loss of energy has to be avoided as much as
possible.
Newly developed electronic components for implementation of switches, such as
modern MOSFETs, support design of highly efficient power supplies. Although fully
integrated solutions are available, applications with external switching stages arewidely used due to flexibility and cost reasons.

2 DC-to-DC conversion methods
2.1 DC-to-DC down-converter
Figure 6 shows the circuit diagram of a simple DC-to-DC down-converter. In
contrast to a linear regulator, this circuit would have 100 % efficiency in case ofideal components application. In practice, there are losses in switching transistor
because the on-resistance is not equal to 0 Ω and also because transistor needs
switching time, which introduces switching losses. Other components add losses
too. Inductor has an ohmic resistance from the wire of the windings and magnetic
core adds losses too. Magnetic core losses result from the change of the magnetic
field which causes motion of small magnetic domains. The bigger the hysteresis of
the core material, the bigger are these losses. Eddy currents cause further loss inthe magnetic core of an inductor. Changing magnetic fields can induce circulating
loops of current which heat up the ferromagnetic material. For high frequency414

switching, the current in the wire no longer uses the whole cross-section, instead it
concentrates closer to the surface. This is a well-known skin effect which leads tohigher ohmic losses.
Also the output capacitor has a residual resistance that leads to energy losses and atemperature increase. Finally, the diode introduces forward voltage losses andreverse current losses. These mechanisms and facts reduce the energy efficiency ofDC-to-DC converters from 75 % to 98 % in real life conditions.

Figure 6 | Simple DC-to-DC down-converter

152Designing in MOSFETs
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MOSFETsMOSFET & GaN FET Application Handbook

The P-channel FET Q1 works as high-side switch. When the FET is switched on, thecurrent in the inductor L1 increases with a linear Δ IL = (ton/L1) x (VIN • VOUT),
curve VOUT is constant.
When the switch is opened, the current continuously flows via the path of thediode D1. The cathode of D1 is negative with the forward voltage VF against
ground. The current decreases linearly. C2 buffers the output voltage. The bigger it
is, the smaller the ripple will be.
Figure 7 depicts a SPICE simulation. The high-side switch is implemented with
NX2301P P-channel FET. It works at the voltage supply V1. The inductance of theinductor is chosen to 68 μH, the output voltage is filtered with 10 μF capacitor.
PMEG2010AEH Schottky diode is selected as a free-wheeling diode. To control
NX2301P, a N-channel driver FET is implemented, which is switched from a squarewave generator with 3.3 V high level (V2). In this example, the switching frequency
is 100 kHz. A load resistor of 10 Ω is connected to the output.

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Figure 7 | SPICE simulation diagram for a simple DC-to-DC down-converter using NX2301P as
high-side switch and PMEG2010AEH as low-side Schottky diode.

Figure 8 shows simulation result. The current IL1 which flows through the inductor
shows a linear increase while Q1 is switched on. The voltage at SW node VSW nearly
equals to the input voltage. When Q1 is switched off, the current through theinductor decreases. The signal SW changes to a negative voltage of about 300 mV,
which is the forward voltage of the Schottky diode. The output current is theaverage of the triangle shaped waveform and is about 330 mA. The output voltage
VOUT is stable at roughly 3.25 V.
In the abovementioned example, the current flows through the inductance for thewhole period of the switching cycle. This mode is called continuous mode of aDC-to-DC converter. Below is a calculation of output voltage. The voltage at an
inductor is:
VL = L x (dIL / dt)					[6]

orVL = L x (ΔIL / Δt)					
		
so

[7]

ΔIL = VL / L x Δt					

[8]The stored energy in an inductor is:416

E = L / 2 x I2					

[9]For the Stationary mode while switch is closed, the energy increase in the inductor
must be identical to the energy loss while switch is open.
Neglecting RDS(on) losses in the switch and the forward voltage of the diode, we get
the formula for ∆IL:

ΔIL = (VIN – VOUT ) × ton = VOUT × toff				[10]VOUT ⁄ VIN= ton ⁄ (ton + toff )= ton ⁄ T			[11]

where T is cycle time and the duty cycle D is:
D = ton ⁄ T						[12]VOUT = VIN × D					[13]In our example:
VOUT = 4,5 V × (7,2 ⁄ 10) = 3,24 V			[14]

If the duty cycle is 1 as a corner case, the switch is always closed and the output
voltage equals the input voltage. If the duty cycle is smaller than 1, the output
voltage is reduced by factor D.
The ripple of the current is:
ΔIL = (VIN – VOUT ) ⁄ L × ton

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				[15]

In our example:
ΔIL = (4,5 V – 3,24 V) ⁄ 68 μH × 7,2 μs=133 mA

		[16]

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4.5

400

(1)
(2)

3.4

2.3

352

304

(3)

1.2

256

0.1

208

-1

0

VSW

10

VOUT

20

30

40

50

160

IL1

Figure 8 | Curves of the current through L 1, SW node voltage and output voltage

If the load current is increased in the continuous mode, the output voltage stays
constant (ideal components) . This means that duty cycle for the switch does notneed to be changed to a significant extent from a DC-to-DC controller IC as long as
the converter runs in the continuous mode. There is a current limit where thecontinuous mode is left. A relevant equation is below:
ΔIL = 2 x IL(average) = 2 x Iload					[17]If curve 3 in Figure 8 moves down by decreasing the output current until the x-axis
is touched, the limit of the continuous mode is reached. From this point onward the
duty cycle has to be reduced in order to keep the same output voltage.
If non-continuous mode is reached, the voltage curve of SW node of the circuitchanges significantly. Normally there is roughly a square wave between VIN and −VF.
If the current through the inductance reaches 0 A, the voltage at the diode changes
from forward direction to reverse. The diode blocks the output capacitor, which is
charged to VOUT from being discharged via L1 (Q1 is still closed). After the current
through L1 went down to zero, SW node shows an oscillation supported by theresonance circuit of L1 and COUT. Figure 9 shows this typical behavior. Circuit on
Figure 7 was modified by reducing the inductance of L1 to 6.8 μH for thisexperiment. This leads to a higher current ripple, and a non-continuous mode.

418

(1)

4

1.5
(2)

0.5
(3)

-2

-4

10

VSW

VOUT

15

20

-0.5

IL1

Figure 9 | Converter in a non-continuous condition, current curve of IL reaches thex-axis at 0 A, curve of SW node jumps from −VF to an oscillation around VOUT

Figure 10 shows a change in the down-converter topology to improve efficiency ofthe simple circuit. The Schottky diode generates forward voltage losses for thetime period when high-side switch is opened. A MOSFET can replace a diode. The
low-side switch needs to be turned on when the upper FET is switched off. The
controller has to take care that there is never an overlap of the on-states of bothtransistors in this case the switching stage would create a short circuit with asignificant current peak, high losses and risk to damage the FETs. Because every
MOSFET contains a body-diode from the source to the drain, the circuit would in
principle work even if Q2 is never switched on. In this case, the body-diode of Q2
would work like a Schottky diode in the simple topology on Figure 6.
Therefore the turn-on time of Q2 is not very critical. If Q2 switches on after Q1 isclosed, the body diode conducts the current from L1.

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Figure 10 | Synchronous DC-to-DC down-converter (FETs including body-diodes)

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2.2 DC-to-DC up-converter
In the previous chapter, inductor-based DC-to-DC down conversion was discussed.
With a small change in the topology, the down-converter can be changed into an
up-converter. Figure 11 shows the topology of a simple DC-to-DC up-converter. If
the low-side switch FET Q1 is closed, the current in the inductance increases:
ΔIL = VIN × ton 						[18]The diode D1 is driven in reverse mode because the anode is connected to theground and the cathode is connected to the positive voltage VOUT at C2. If theswitch is closed, the current IL continues to flow through D1 into the output. If theconverter runs in a stationary mode, we can calculate:
ΔIL= VIN ⁄ L × ton= (VOUT – VIN ) ⁄ L × toff				[19]VIN × ton= (VOUT – VIN ) × toff				[20]

VOUT = VIN × (ton / toff + 1)					[21]where the duty cycle is:
D = ton ⁄ T						[22]

T= ton + toff						[23]		VOUT = VIN × (ton + toff ) / toff = VIN x T / (T - ton)
= VIN x 1 / (1 - ton / T)						[24]= VIN x 1 / (1 - D)
The corner cases of the equation show that for D = 0, which means that thetransistor is never switched on, VOUT = VIN. It makes sense to consider losslesscomponents. Lossless means a diode with no forward voltage and an inductancewithout an ohmic resistance of the windings and the additional loss mechanisms
discussed in the previous chapter. If D gets close to 1, the output voltage increases
rapidly. This is critical for safe operation because high duty cycle can result in very
high voltages at the FETs drain.

420

Figure 11 | Simple DC-to-DC up-converter

Figure 12 shows SPICE simulation. The low-side switch is implemented with
PMV20XN N-channel MOSFET in SOT23 package and PMEG2010AEH Schottky
diode. The converter is switched with 100 kHz signal control signal with a duty cycleof 0.5.

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Figure 12 | SPICE simulation diagram for a simple DC-to-DC up-converter example with PMV20XN
N-channel FET and PMEG2010AEH Schottky diodeFigure 13 shows simulation results. Curve 2 represents output voltage. For ideal
components, the output voltage would be twice as high as the input due to theduty cycle of 0.5. In practice, the forward voltage of the diode reduces the output
voltage. Curve 1 shows drain voltage VD of the N-channel FET. It switches between
ground level and VD(max) and equals:
VD(max) = VIN × 1 ⁄ (1 – D ) + VF 		.

			[24]421

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In the simulated case with the duty cycle D = 0.5,
VD(max) = 2 × VIN + VF					[25]

VSW

VOUT

IL1

Figure 13 | Simulation results, simple up-converter

Similar to DC-to-DC down-converter, energy efficiency of the up-converter also canbe improved if the Schottky diode is replaced by a FET, which switches on for thecorrect phase in the switching cycle. Figure 14 shows the topology of synchronous
DC-to-DC up-converter.

Figure 14 | Synchronous DC-to-DC down-converter

2.2.1 DC-to-DC up- and down-converter
If two topologies of the DC-to-DC down-converter and up-converter are combined
as shown on Figure 15, the output voltage can either be reduced or boosted inrelation to the input voltage. The MOSFET switches need to be controlled in aproper way to allow the conversion in both directions. Q3 and Q4 can switch422

similarly to the DC-to-DC down-converter shown on Figure 10. In addition, Q2 must
be switched on constantly to connect the inductor to the output capacitor. For the
up-conversion mode, MOSFETs Q2 and Q1 work as the switching stage, as
described for the synchronous up-converter on Figure 14. The MOSFET Q3 isconstantly switched on to connect the inductor to the input supply voltage in thiscase.

Figure 15 | Up-down converter

3 Medium power DC-to-DC down-converter
using small-signal MOSFETs

15
2
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MOSFETsMOSFET & GaN FET Application Handbook

3.1 DC-to-DC down-converter application
board
Figure 16 shows an application Printed-Circuit Board (PCB) with Nexperia smallsignal MOSFETs implemented in a DC-to-DC step-down converter. Nexperia offers
small-signal MOSFETs in small SMD packages such as SOT457, SOT23, SOT223 andDFN2020MD-6 (SOT1220). Many of these MOSFETs provide very low RDS(on)together with a good switching performance.
The topology of the application board on Figure 16 is a synchronous downconverter same as in Section 2.3.1. The circuit contains a controller LTC3851 ofLinear Technology Corporation. Two N-channel MOSFETs build switching stage. The
high-side switch connects the node with the inductor to the input supply.
Therefore, it is necessary to have a control voltage available that is higher than the
input voltage itself. This extra voltage for the control of the gate of the upper
MOSFET is generated with a charge pump. The capacitor C25 is connected to the
SW node, the switched output and via Schottky diode to a stabilized voltage

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INTVCC (pin 12). INTVCC is provided by an internal 5 V LDO. The capacitor ischarged via the diode when the low-side switch is turned on. In this case, one sideof C25 is connected to ground. If Q2 is turned off and Q1 is switched on, thecharged capacitor gets connected to VIN. At the pin BOOST (pin 14) a voltage of
VIN + INTVCC − VF (forward voltage of the diode) can be measured. This boosted
voltage can drive the high-side switch properly. Low current Schottky diodes are
sufficient for the charge pump (for example, BAT54J, 1PS76SB40 or 1PS76SB21).
These diodes are supplied in the SMD packages like SOD323F and SOD323.
The LTC3851 controller contains a 0.8 V precision reference voltage for the output
voltage regulation. The output of the down-converter is fed back to the pin FB. A
resistor divider formed by R41 + R39 and R38 adjusts the output voltage.The
equation for the output voltage is:
VOUT = 0,8 V × (1 + (R41 + R39) ⁄ R38 ) 		[26]

The controller works with a constant frequency. As described in Section 2.3.1, theoutput voltage of DC-to-DC down-converter can be controlled rather easily forhigher currents, but low current conditions are more ambitious for the control. The
duty cycle needs to be changed significantly or the controller can change to adifferent control mode like burst operation. For the LTC3851 there are three
options: forced continuous operation, burst mode operation and a pulse-skip
mode.
Burst mode operation gives better efficiency, but more ripple and a higher
ElectroMagnetic Interference (EMI) level. The best mode depends on thespecification and requirements of the end application.
The switching frequency can be programmed in a range from 250 kHz to 750 kHz.
The resistor R30 determines the frequency. Alternatively the controller cansynchronize the internal oscillator to an external clock source (MODE/PLLIN, pin 1).
In this mode an RC network needs to be connected to pin 2 (FREQ), which serves as
PLL loop filter.

424

49k9

1
2
3
4

INTVCC

R41

43k2

GND
ILIM

SENSE+BGFB
SENSE-INTVCC

VIN

ITH

TK/SS

BOOST

TG

FREQ
RUN

SW

MODE

LTC3851EPN

10

11

12

13

14

15

16

100n

GND

PMEG3010ER

PMPB20EN

+

+

C18

IHL-5050EZ-01-Coil

+

+

1
2
3

1
2
3

GND_OUTGND_IN

Designing in MOSFETs
DC-to-DCfor conversion
safe and reliablewith small-signal
gate-drive operation
MOSFETs

Figure 16 | Circuit diagram for reference application of DC-to-DC down-converter with Nexperia small-signal MOSFET PMN15UN and LTC3851 as controller

100n

100n

GND

GND

MODE

MOSFET & GaN FET Application Handbook

15
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3.2 Small-signal MOSFETs suitable for
DC-to-DC conversion
DC-to-DC converters can be found in many applications. Topology of step-down
converter controller with an external FET stage are often implemented incomputing and consumer applications. Modern concepts with latest generation
SOC (System-on-Chip) solutions require many separate supply voltages. These canbe processors on motherboards, in notebooks and tablet PCs, on core chips of
LCD-TV or a set-top box.
The power requirements range from more than hundred watts down to a few
watts. In desktop PCs, DC-to-DC converter can be found on the motherboard which
provide a current capability of up to 100 A and an output power of up to 130 W.
MOSFETs in switching stages are Loss-Free Package (LFPAK) types and to a growing
extent Quad Flat-pack No-lead (QFN) 5 × 6 packages. For net- and notebooks thepower requirements are smaller. The power consumption ranges from 18 W to
55 W. The switching MOSFETs are mainly SO-8 and QFN 3 × 3 types. In consumer
applications such as LCD-TVs and set-top boxes as well as in low-power netbooks ortablet PCs, power requirements from 7 W to 15 W can be found.
For medium power range small-signal MOSFETs can replace SO-8 versions
nowadays in smaller packages like QFN 3 × 3, but also in QFN 2 × 2 or SOT457.

3.3 Dimensioning aspects for the inductor andoutput capacitor
In order to reach a desired current ripple, choose carefully inductance of theinductor used in the down-converter. With a bigger current ripple, the output
voltage shows a larger ripple. The ripple increases the smaller the inductancebecomes and the higher the input voltage is. Furthermore it increases if theswitching frequency is reduced.
ΔIL can be calculated:
ΔIL = VIN ⁄ L × ton= VOUT ⁄ L × toff			

[27]

with:
T = ton + toff = 1 ⁄ f		

			

[28]

we get:ΔIL= (VOUT ⁄ L ) × (1 – VOUT ⁄ VIN ) × 1 ⁄ f 				[29]426this means:
L = (VOUT ⁄ ΔIL ) × (1 – VOUT ⁄ VIN ) × 1 ⁄ f				[30]For the corner case in which circuit runs at the limit of the continuous mode,current goes down exactly to zero before it increases again and we get simple
relation:
ΔIL = 2 × Iaverage					[31]Putting equation 32 into the formula for ΔIL:
L = VOUT × (1 – VOUT ⁄ VIN ) ⁄ 2 × Iaverage × f 		[32]

In practice, the ripple current ΔIL is about 30 % of the maximum current, as a rule ofthumb.
The ripple of the output voltage does not depend on the chosen inductance andthe ∆IL only, but also on the capacitance of the output capacitor. The bigger thecapacitor, the smaller the ripple is. Figure 17 shows the waveform of the current
into the capacitor. For a lossless capacitor, there is basic equation:t2

Vc = 1 × ∫ Ic × dt						[33]
C tolc

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t
t0

t1

ton

ΔIL

t2

toﬀ

Figure 17 | Capacitor current IC versus time

For t0 to t1:
IC = ΔIL ⁄ ton × t					[34]427Designing in MOSFETs
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and for t1 to t2
IC = ΔIL ⁄ toff × t						[35]the integral formula for the capacitor ripple voltage can be written as:															[36]With:
T = ton + toff = 1 ⁄ f 						[37]		
the result is:
VC _ripple = ΔIL ⁄ (C × 8 × f)					[38]For real capacitors take into account an Equivalent Series Resistance (ESR). So we
get an equation:
VC _ripple = ΔIL × (ESR + 1 ⁄ (8 × f × Cout ) )				[39]3.4 MOSFET losses calculation
For MOSFETs used as switches consider two loss processes. One is the ohmic loss
caused by the residual on-state resistor RDS(on). The second loss process happens at
the switching transients. Because FETs are not ideal switches that can change fromoff- to on-state or reverse without a small turn-over time.
The RDS(on) losses are also called I2R-losses and they can be calculated:
Pup _side _switch = D × (IOUT)2 × (1 + δ ) × RDS(on) = VOUT / VIN x (IOUT)2 x (1 + δ) × RDS(on) [40]with duty cycle:
D = ton ⁄ T							[41]The term 1+δ contains the temperature dependency of RDS(on) of a MOSFET. δ has
typically a value of:δ = (0, 005 ⁄ °C) × (Tj – 25°C) 				[42]For the low side switch, there is a similar formula. Because synchronous FET is428

conducting while the high-side switch is closed, the I2R looses can be calculatedwith the equation:
Plow _side _switch = (1 – D ) × (IOUT)2 × (1 + δ) × RDS(on) =1 – (VOUT ⁄ VIN ) × (IOUT)2 × (1 + δ ) × RDS(on)			 [43]Regarding the transition losses, only the high-side switch suffers from thismechanism. The reason is that the implemented free-wheeling diode (D1 on Figure16) is getting conductive. It reduces the voltage over the synchronous FET to its
small forward voltage VF. If the circuit does not contain a free-wheeling diode, thesituation is different: losses of the body diode need to be added to the RDS(on)losses of the FET. In general, efficiency suffers from the higher VF and reverse
recovery time of the body diode if there is no free-wheeling Schottky diodeimplemented.

15
2
Designing in MOSFETs
DC-to-DCfor conversion
safe and reliablewith small-signal
gate-drive operation
MOSFETsMOSFET & GaN FET Application Handbook

Figure 18 | Diagram for switching behavior tests for MOSFET

Figure 18 shows a test circuit for the switching behavior of a MOSFET. The parasitic
capacitors from the gate to the source CGS and from the drain to the gate CDG aredepicted explicitly. Current source IG controlls gate. At the drain another current
source is connected towards VSS with a free-wheeling diode in parallel. As long as
the FET is closed, the current flows through this diode.
Figure 19 shows how the switching-on process looks like. If the current source IG is
switched on, voltage at CGS rises with a linear curve until gate-source thresholdvoltage VGS(th) is reached. At this time a drain current starts to flow. This means that
the FET remains in the off-state during the time period t0.
During t1 the drain current increases. Also the gate voltage increases until VGS(pl) is
reached. VGS(pl) is commonly known as plateau voltage of a MOSFET. It is normallynot explicitly mentioned in data sheets, but it can be derived from the diagram

429

Designing in MOSFETs
DC-to-DCfor conversion
safe and reliablewith small-signal
gate-drive operation
MOSFETs15
2

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gate charge versus gate-source voltage which can be found in detailed data sheets.
After the time period t0 and t1, the charge is Q0 = Vpl × (CGS + CDS ) .
In the next time period t2, the drain voltage decreases and gate-source voltage VGS
stays constant at VGS(pl). CDS gets charged in the reverse direction with the charge
Q1 which is:
Q1 = VSS × CDS[44]

CDS is similar to the Miller capacitance known from bipolar transistors and has asignificant impact on the switching performance of a MOSFET.
During t3 the gate voltage increases again until the current source is stopped
where the desired maximum gate voltage is reached. The RDS(on) of the FET is
reduced further. The gate driver provides an additional charge Q2, which is:
Q2 = (VGS(t4) – VGS(pl) ) × (CGS + CDS)

[45]

The total charge follows the equation: QG = Q0 + Q1 + Q2
This charge can easily exceed 100 nC for a power MOSFET. With the equation:
IG = QG ⁄ ts

[46]

the gate current can be calculated to achieve a switching time ts. Therefore, if smalltransition times are desired, in order to keep the switching losses small, apply
powerful drivers to control MOSFETs.

VGS
VGS(pl)VGS(th)

Q0

Q1

Q2

t

VDS
Vin

ID
t0

t1

t2

t3

t4

t

Figure 19 | Turn-on process for MOSFET, VGS, VDS and ID curves

During the time t1, there is the full input voltage at the FET while drain current ID
increases. In the next time section t2, ID is constant while drain-source voltage VDS

430

decreases. The major switching losses occur during these two time periods in theswitching process. Rather small losses during t3 are neglected. During t3 RDS(on)falls to the minimum value that is reached when the final VGS voltage is reached.
Switching losses during turn-on occur in the time period t1 and t2. The mostdominant time is t2 where the gate voltage of the MOSFET remains at the plateau
voltage V(pl). The losses can be calculated as:

15
2
Designing in MOSFETs
DC-to-DCfor conversion
safe and reliablewith small-signal
gate-drive operation
MOSFETsMOSFET & GaN FET Application Handbook

PSW(on) = VIN × I ⁄ 2 × (t3 + t1) × 1 ⁄ T		
[47]
with the switching frequency of converter: fSW = 1 ⁄ T
Turn-off behavior of a MOSFET is similar to the turn-on process. Total switchinglosses can be summarized as:

PSW = VIN × 1 ⁄ T × (Imin ⁄ 2 × ton + Imax ⁄ 2 × toff )		
[48]
Switching time depends on the current drive capabilities of the driver device andthe gate resistance of the FET. If we assume an identical drive current for turn-on
and turn-off event, switching time equals: tSW = QG ⁄ Idrive
For the LTC3851 tSW can be estimated roughly as follows. Rdrive is about 2 Ω for thecontroller. The relevant voltage is driver voltage INTVCC − V(th), so we get:tSW = QG × Rdrive ⁄ (Vdrive – VGS(th))

431

Designing in MOSFETs
DC-to-DCfor conversion
safe and reliablewith small-signal
gate-drive operation
MOSFETs152432

Nexperia | A Power Design Engineer's Guide

4 Figures
Figure 6. Simple DC-to-DC down-converter ............................................................... 415
Figure 7. SPICE simulation diagram for a simple DC-to-DC down-converter
using NX2301P as high-side switch and PMEG2010AEH as
low-side Schottky diode................................................................................ 416
Figure 8. Curves of the current through L 1, SW node voltage and output
voltage ............................................................................................................. 418
Figure 9. Converter in a non-continuous condition, current curve of IL
reaches the x-axis at 0 A, curve of SW node jumps from −VF
to an oscillation around VOUT........................................................................ 419
Figure 10. Synchronous DC-to-DC down-converter (FETs including
body-diodes) ................................................................................................... 419
Figure 11. Simple DC-to-DC up-converter .................................................................... 421
Figure 12. SPICE simulation diagram for a simple DC-to-DC up-converter
example with PMV20XN N-channel FET and PMEG2010AEH
Schottky diode................................................................................................ 421
Figure 13. Simulation results, simple up-converter..................................................... 422
Figure 14. Synchronous DC-to-DC down-converter .................................................... 422
Figure 15. Up-down converter........................................................................................ 423
Figure 16. Circuit diagram for reference application of DC-to-DC
down-converter with Nexperia small-signal MOSFETPMN15UN and LTC3851 as controller........................................................ 425
Figure 17. Capacitor current IC versus time................................................................... 427
Figure 18. Diagram for switching behavior tests for MOSFET.................................. 429
Figure 19. Turn-on process for MOSFET, VGS, VDS and ID curves................................ 430

Chapter 16
Load switches formobile andcomputing
Application Note: AN90017

Load switches for mobile and computing

16

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Load switches for mobile and computing

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1 Types of load switches
This chapter presents simple load switches in mobile and computing applications
There are several reasons why a circuit or subsystem is required to be disconnectedfrom the power supply by means of a load switch. A very simple and very common
reason is, that it helps saving power. An unpowered subsystem may not consume
power due to leakage or standby currents. In portable electronic devices, load
switches can be used to prevent damage from electrical surges, incorrect battery
insertion, and other damaging events that can enter through the power source.
Before diving deeper into the key parameters, let’s look at the different types ofload switches, first. A high-side load switch connects or disconnects a power source
to a load. The switch is controlled by an external enable signal. High-side switchessource current to a load. Low-side switches connect or disconnect the load toground, thus sinking current from the load.
A load switch can be easily implemented with a MOSFET pass transistor. The
MOSFET passes current from the power source to the load and is turned on or off
via a control signal. Providing the control signal to the MOSFET, a gate-drive circuit
connects to the MOSFET’s gate to switch the MOSFET on or off.

1.1 P-channel MOSFET load switch
Using a P-channel MOSFET a high-side load switch can be implemented byconnecting the input voltage to the MOSFET source and the load to the MOSFET
drain, see Figure 1. Pulling the gate low will enable the current flowing into theload.

P-Ch MOSFET

Figure 1 | P-channel MOSFET load switch434

1.2 N-channel MOSFET load switches
N-channel MOSFETs have lower ON resistance values than P-channel devices of thesame size. However, to get the lower resistance values, a high voltage is needed todrive a MOSFET’s gate in the implementation of a high-side load switch using an
N-channel MOSFET, see Figure 2. A voltage higher than the input voltage Vin has tobe provided for proper gate drive otherwise the MOSFET would not turn on
entirely but run in linear mode with high power losses.

N-Ch MOSFET

16
Load switches for mobile and computing

MOSFET & GaN FET Application Handbook

Figure 2 | N-channel MOSFET load switch with high voltage control line

If a high enough voltage to drive the gate is not available, a charge-pump circuit canbe used to increase the drive voltage applied to the MOSFET's gate, see Figure 3.
While this increases the complexity of the circuit, it is rewarded with the lower ON
resistance of the N-Channel MOSFET. However a charge-pump circuit will dissipate
some power so that in very energy critical applications that may stay most of thetime in stand-by mode, a P-Channel topology can be more efficient.

N-Channel MOSFET

Figure 3 | N-channel MOSFET load switch with high voltage control line

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Without high voltage or additional circuitry, the N-channel MOSFET can be used in a
low-side load switch. This implementation of a low side load switch is shown in
Figure 4. A downside of the low-side load switch is the slight lifting of the ground
potential of the load. This needs to be considered especially if the load has
communication lines with external components.

N-Channel MOSFET

Figure 4 | N-channel MOSFET low-side load switch

More advanced load switch features include reverse voltage protection, reverse
current protection. This can be implemented with MOSFETs in common-drain orcommon-source configuration as shown in the next section

1.3 Reverse current protected load switches
Load switches based on a single MOSFET topology can block current in one
direction only. As MOSFETs have an inherent body diode they act like a diode in
off-state if driven reverse. In some applications it is required to be able to blockcurrent flow in both directions.
Applications, that need blocking of reverse currents are e.g battery driven
applications, where discharge of the battery shall be prevented for fault conditions
like a short at the charger connector side or in case of electrical failure causing
leakage in connected cables and AC adapters.
A common way to block reverse currents is by applying diodes. However, thisfunction can be realized far more efficiently with MOSFET load switches. In order toachieve blocking for both current directions, two MOSFETs have to be put in series
with reverse polarity to each other. In this case one of the body diodes can blockcurrent flow if not both FETs are turned on. This approach allows to create twoalternative topologies of back-drive protected load switches. Either the drain
contacts or the source contacts can be connected.

436

P-Channel MOSFETs

Figure 5 | Reverse current protected common drain load-switch

Control of the back-drive protected load switches can be done with separate gate
drive or connected gates. The common drain topology, see Figure 5 has theadvantage for a semiconductor component that the drain contact is locatedtowards the substrate side. So, the common drain can be attached directly to thelead frame with a good thermal path into the PCB. For mobile applications often
CSP packages can be found. Here the source and gate contacts are facing down to
the PCB whereas a drain contact needs an additional down contact. If there is noneed to make the drain accessible necessarily. In this case a CSP design becomesvery space efficient because no chip area needs to be sacrificed for down
connections of drain.

RDS(on)Load switches for mobile and computing

16

MOSFET & GaN FET Application Handbook

Back Metal
Chip 145um

Solder ballFigure 6 | Turn-on resistance (RDS(on)) of a single CSP MOSFET

437

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Nexperia | A Power Design Engineer's Guide

Back metalISS(on)Chip 100-200um
S1

Polymide

S2

Bump Cu/Sn
Figure 7 | Turn-on resistance (RDS(on)) of a common-drain CSP MOSFETThe common source topology requires an access to the source connection between
the MOSFETs. The gate has to be discharged after a turn on, see Figure 8. This does
not work without access to this node if the common sources might float.

P-Channel MOSFETs

Figure 8 | Reverse current protected common source load-switch

1.4 Reverse polarity protection and supply
OR-ing
In cases where no switching of the load is needed a single MOSFET can be used forreverse polarity protection only, Figure 9. The MOSFET can replace simple diodes
e.g. in supply OR-ing applications this has the advantage of a lower voltage drop
across the MOSFET compared to the voltage drop across the diode.

438

P-Channel MOSFET

Figure 9 | Reverse current and reverse polarity protection with a P-Ch MOSFET

Another important application of back-drive load-switches is power OR-ing, see
Figure 10. If a circuit is able to be powered by one of several power supplies thencurrent flow back into a non-selected supply has to be avoided.

Load switches for mobile and computing

16

MOSFET & GaN FET Application Handbook

P-Channel MOSFETs
Vin1

internal supply

Vin2

Or-ing / supply
priotity circuit

Figure 10 | Power supply OR-ing with P-Channel MOSFETs

2 Important parameters
Important key parameters of a load switch are the turn-on resistance (RDS(on)) of the
MOSFET that connects between the voltage input and voltage output pins, themaximum current ID(max) that the transistor withstand, and the maximum voltage
VDS(max) that the circuit can withstand. The lower the turn-on resistance, the lower
the power dissipation of the transistor and the lower the voltage drop from input
to output.

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Load switches for mobile and computing

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Nexperia | A Power Design Engineer's Guide

While RDS(on), ID(max), and VDS(max) are parameters of the MOSFET, the maximum
voltage drop and the maximum power dissipation of a loadswitch can be calculate
for a given current I with the formulas below.
Vdrop = I * RDS(on)(max
P = I * Vdrop = I2 * RDS(on)(max
Today’s MOSFETs typically have turn-on resistance values in the tens of milliohms,
so, for example, if the load switch has an ON resistance of 50 milliohms andcontrols a 200 mA load, the MOSFET dissipates just 2 mW when ON, and has an
input-to-output voltage drop of 10 mV. Even a peak current of 1 A would only causea voltage drop of 50 mV and peak power dissipation of 50 mW.
As load switch circuits are active whenever the power is on and are thus designed
to have low leakage currents, see Nexperia Application note AN90009 (chapter 14
in this book) - Leakage of small-signal MOSFETs.

3 Figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.

P-Channel MOSFET load switch...................................................................434
N-Channel MOSFET load switch with high voltage control line.............435
N-Channel MOSFET load switch with high voltage control line ............435
N-Channel MOSFET low-side load switch...................................................436
Reverse current protected common drain load-switch...........................437
Turn-on resistance (RDS(on)) of a single
Chip-scale package MOSFET ........................................................................437
Figure 7. Turn-on resistance (RDS(on)) of a common-drain
Chip-scale package MOSFET.........................................................................438
Figure 8. Reverse current protected common source load-switch........................438
Figure 9. Reverse current and reverse polarity protection with a
P-Ch MOSFET...................................................................................................439
Figure 10. Power supply OR-ing with P-Channel MOSFETs........................................439

16
Load switches for mobile and computing

MOSFET & GaN FET Application Handbook

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Level shifting techniques in I2C-bus design

17

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Nexperia | A Power Design Engineer's Guide

Chapter 17
Level shifting
techniques in
I2C-bus design
Application Note: AN10441

Level shifting techniques in I2C-bus design

17

MOSFET & GaN FET Application Handbook443

Level shifting techniques in I2C-bus design

17

Nexperia | A Power Design Engineer's Guide

1 Introduction
Logic level shifting may be required when interfacing legacy devices with newer
devices that use a smaller geometry process. For bidirectional bus systems like the
I²C-bus, such a level shifter must also be bidirectional, without the need of adirection control signal. The simplest way to solve this problem is by connecting a
discrete MOSFET to each bus line.
Present technology processes for integrated circuits with clearances of 0.5 µm andless limit the maximum supply voltage and consequently the logic levels for thedigital I/O signals. To interface these lower voltage circuits with existing 5 V
devices, a level shifter is needed. For bidirectional bus systems like the I2C-bus, sucha level shifter must also be bidirectional, without the need of a direction control
signal. The simplest way to solve this problem is by connecting a discrete MOSFET
to each bus line.

2 Bidirectional level shifter for Fast-mode andStandard-mode I2C-bus systems
In spite of its surprising simplicity, such a solution not only fulfils the requirementof bidirectional level shifting without a direction control signal, it also:• Isolates a powered-down bus section from the rest of the bus system• Protects the ‘lower voltage’ side against high voltage spikes from the‘higher-voltage’ side.
The bidirectional level shifter can be used for both Standard-mode (up to100 kbit/s)or in Fast-mode (up to 400 kbit/s) I2C-bus systems. It is not intended for Hs-mode
systems, which may have a bridge with a level shifting possibility.

2.1 Connecting devices with different logic
levelsDifferent voltage devices could be connected to the same bus by using pull-up
resistors to the supply voltage line. Although this is the simplest solution, the lower
voltage devices must be 5 V tolerant, which can make them more expensive tomanufacture. By using a bidirectional level shifter, however, it is possible tointerconnect two sections of an I2C-bus system, with each section having a different
supply voltage and different logic levels. Such a configuration is shown in
Figure 1. The left ‘low-voltage’ section has pull-up resistors and devices connected444

to a 3.3 V supply voltage; the right ‘high-voltage’ section has pull-up resistors anddevices connected to a 5 V supply voltage. The devices of each section have I/Os
with supply voltage related logic input levels and an open-drain output
configuration.
The level shifter for each bus line is identical and consists of one discrete N-channel
enhancement MOSFET; TR1 for the serial data line SDA and TR2 for the serial clock
line SCL. The gates (g) have to be connected to the lowest supply voltage VDD1, thesources (s) to the bus lines of the ‘lower-voltage’ section, and the drains (d) to thebus lines of the ‘higher-voltage’ section. Many MOSFETs have the substrateinternally connected with its source, if this is not the case, an external connection
should be made. Each MOSFET has an integral diode (n-p junction) between thedrain and substrate.
VDD2 = 5 V

VDD1 = 3.3 V
Rp
SDA1

g

Rp

TR1

s

Rp

Rp
SDA2

d
g TR2

SCL1

3.3 V DEVICEs

3.3 V DEVICE

Level shifting techniques in I2C-bus design

17

MOSFET & GaN FET Application Handbook

SCL2

d

5 V DEVICE

5 V DEVICE

mgk879Figure 1 | Bidirectional level shifter circuit connecting two different voltage sections in an I2C-bus
system2.1.1 Operation of the level shifter
The following three states should be considered during the operation of the level
shifter:
1. No device is pulling down the bus line. The bus line of the ‘lower-voltage’section is pulled up by its pull-up resistors Rp to 3.3 V. The gate and thesource of the MOSFET are both at 3.3 V, so its VGS is below the thresholdvoltage and the MOSFET is not conducting. This allows the bus line at the‘higher-voltage’ section to be pulled up by its pull-up resistor Rp to 5 V. So
the bus lines of both sections are HIGH, but at a different voltage level.
2. A 3.3 V device pulls down the bus line to a LOW level. The source of the
MOSFET also becomes LOW, while the gate stays at 3.3 V. VGS rises above

445

Level shifting techniques in I2C-bus design

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Nexperia | A Power Design Engineer's Guide

the threshold and the MOSFET starts to conduct. The bus line of the 		‘higher-voltage’ section is then also pulled down to a LOW level by the 3.3 V
device via the conducting MOSFET. So the bus lines of both sections go LOW
to the same voltage level.
3. A 5 V device pulls down the bus line to a LOW level. The drain-substrate
diode of the MOSFET the ‘lower-voltage’ section is pulled down until VGS
passes the threshold and the MOSFET starts to conduct. The bus line of the‘lower-voltage’ section is then further pulled down to a LOW level by the 5 V
device via the conducting MOSFET. So the bus lines of both sections go LOW
to the same voltage level.
The three states show that the logic levels are transferred in both directions of thebus system, independent of the driving section. State 1 performs the level shiftfunction. States 2 and 3 perform a ‘wired-AND’ function between the bus lines ofboth sections as required by the I2C-bus specification.
Supply voltages other than 3.3 V for VDD1 and 5 V for VDD2 can also be applied, e.g.,
2 V for VDD1 and 10 V for VDD2 is feasible. In normal operation VDD2 must be equal toor higher than VDD1 (VDD2 is allowed to fall below VDD1 during switching power on/off).

3 Figures
Figure 1. Bidirectional level shifter circuit connecting two different voltage ............
sections in an I2C-bus system........................................................................ 445

17Level shifting techniques in I2C-bus design

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Understanding Power GaN FET data sheet parameters18448
Nexperia | A Power Design Engineer's Guide

Chapter 18
Understanding
Power GaN FET
data sheet parameters
Application Note: AN90005

Understanding Power GaN FET data sheet parameters

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MOSFET & GaN FET Application Handbook449

Nexperia | A Power Design Engineer's Guide

1 Introduction
This chapter explains the content of Nexperia Power GaN FET data sheets.
Nomenclature, pinning and key parameters are detailed.
The chapter details the data sheet for the GAN063-650WSA GaN FET device from
Nexperia. Visit the product information page on Nexperia.com to download thelatest version of the full data sheet.

2 Nomenclature
The device name (type number) is shown at the top of the data sheet. The namecontains some important information about the device. In the below example for
GAN063-650WSA, the maximum on-state resistance and the limiting drain-source
voltage are specified, together with letter codes indicating the package, themounting base electrical connection and the technology generation.

Nexperia
GaN FETR DS(on) (max.)

Drain-source voltage (max.)
650 V

63 mΩ

{
{

{

Understanding Power GaN FET data sheet parameters18

GAN 063-650WSA

w = package code TO247
S = TAB connected to Source
A = Nexperia technology generation 1

Figure 1 | GaN FET Nomenclature

3 Pinning information
Table 1: Pinning information

450

Pin

SymbolDescription

1

G

gate

2

S

source

3

D

drain

mb

S

mounting base;connected to source

Simplified outline

Graphic symbol

mb

D

G

1

2

3

TO-247 (SOT429)

S

aaa-028116

The Nexperia GAN063-650WSA is packaged in a TO247 (SOT429) package with asource tab. The traditional TO-247 provides excellent heat transfer, and therefore
excellent power-handling capability.
mounting baseconnected to source

aaa-029292

Note: the pin functions are different from a standard MOSFET: pin 1 is the gate,pin 2 is the source and pin 3 is the drain.

G

gate
loop

D

S

G

ZL

power
loop

aaa-029293

Figure 2 | Conventional pin assignment		

gate
loop

S

D

ZL

power
loop

aaa-029294

Understanding Power GaN FET data sheet parameters

18

MOSFET & GaN FET Application Handbook

Figure 3 | Nexperia pin assignment

Using pin 2 as the source pin allows the circuit designer to keep the gate loop andthe power loop separate. This facilitates a very clean design and greatly reduces or
eliminates cross coupling. The PCB design can be optimised to take advantage ofthis feature.

4 Two-chip, integrated, normally-off powerswitchThe full graphic symbol of a GaN FET is shown in the pinning information section of
the data sheet. Note that the symbol includes two devices:D

G
S

aaa-028116

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Understanding Power GaN FET data sheet parameters

18

Nexperia | A Power Design Engineer's Guide• Low-voltage Si MOSFET with a p-n body diode• High-voltage GaN FET without a p-n body diodeThe full graphic symbol shows that the device is a two chip integrated switch.
Functionally the switch is normally off. Internally the device is built with two chips.
The high-voltage GaN HEMT or FET is normally on, which is the type most naturally
made with GaN, and is combined with a high performance normally-off Si MOSFET
specifically developed to complement the GaN HEMT. The two chips are integrated
with absolute minimum inductance between them.

Q2
il

Vi

D2

VIN
VD

Q1

D1

aaa-029295

Figure 4 | Simplified GaN FET symbol

In circuit schematics where just the basic switching function is important, a simple
N-channel FET symbol is used to represent the complete, integrated switch (see Q1circled in red above). When you see the simple symbol, understand that it
represents this two-chip integrated combination.

5 GaN FET limiting values
The limiting values table provides the range of operating conditions allowed for the
GaN FET. The conditions are defined in accordance with the Absolute Maximum
Rating System (IEC 60134).
Operation outside of these conditions is not guaranteed, so it is recommended thatthese values are never exceeded. Doing so risks immediate device failure or
reduced lifetime of the GaN FET. To calculate how the limiting values change withtemperature de-rating curves are provided.
The limiting values table for the GAN063-650WSA is given as an example of astandard limiting values table, in Table 2.

452

Table 2: Limiting values
In accordance with the Absolute Maximum Rating System (IEC 60134).
Symbol Parameter

Conditions

Min

Max

Unit

VDS

drain-source voltage-55 °C ≤ Tj ≤ 175 °C

-

650

V

VTDS

transient drain to source
voltagepulsed; tp = 1 µs; δfactor= 0.01

-

800

V

VGS

gate-source voltage

-20

20

V

Ptot

total power dissipation

Tmb = 25 °C; Figure 8

-

143

W

ID

drain current

VGS = 10 V; Tmb = 25 °C

-

34.5

A

VGS = 10 V; Tmb = 100 °C

-

24.4

A

IDM

peak drain current

pulsed; tp ≤ 10 µs; Tmb = 25ºC -

150

A

Tstg

storage temperature

-55

150

°C

Tj

junction temperature

-55

175

°C

Tsld(M)

peak soldering
temperature

-

260

°C

Source-drain diodeIS

source current

Tmb = 25 °C; VGS = 0 V

-

34.5

A

ISM

peak source current

pulsed; tp ≤ 10 µs; Tmb = 25 °C -150

AUnderstanding Power GaN FET data sheet parameters

18

MOSFET & GaN FET Application Handbook

5.1 Drain source voltage , VDS

This 650 V rating is the maximum value that will give you the desired product life.
GaN FETs do not have an avalanche breakdown mechanism. The Nexperia GaN FET
does not typically show any significant leakage current until a VDS well beyond
800 V is reached. So there is an extra margin in VDS before excess leakage occurs. If
the GaN FET is subjected to a sufficiently high voltage, well beyond the specified
maximum, then because there is no clamping mechanism damage and failure will
occur. Since GaN FETs do not have an avalanche breakdown mechanism they are
immune to cosmic radiation and so no further derating is required.

453

Understanding Power GaN FET data sheet parameters

18

Nexperia | A Power Design Engineer's Guide• VDS is the maximum voltage the device is guaranteed to block between drain andsource terminals in the off-state. VDS is a DC rating• VDS is not limited by avalanche breakdown; the rating can be applied over theentire operating range of -55 °C to 175 °C in contrast to the VDS for a Si MOSFET
which must be de-rated below 25 °C

5.2 Transient drain-source voltage, VTDSVTDS is the Maximum repetitive transient voltage the device is guaranteed to block
between drain and source in the off state. This transient rating, applies over theentire operating temperature range.
Pt2

duty cycle δ =

t1
t2

t1

t

006aaa812

Figure 5 | Duty Cycle Definition
800 V

+
DUT
3 MΩ

VG

VDS

tp

tp ≥ 1 µs

800 V

800 V

900 V
MOSFET

0V

0V

10 tp
pulse width ≥ 1 µs
duty ratio = 0.1

aaa-029296

Figure 6 | VTDS test circuit			454

aaa-029297

Figure 7 | VTDS waveform

5.3 Gate source voltage, VGS and total power
dissipation, Ptot

VGS
Maximum voltage the device is guaranteed to block between the gate and source
terminals. This is a DC rating, and applies over the entire operating temperature
range.
Ptot
Ptot is the Total Power dissipation is the maximum for a device with a mounting
base temperature of 25 °C.
120

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18

MOSFET & GaN FET Application Handbook

Pder
(%)
80

40

0

0

50

100

150

Tmb (°C)

200

03aa16

Figure 8 | Normalised total power dissipation as a function of mounting base temperaturePower dissipation is calculated as that which would take the device to the
maximum allowed junction temperature while keeping the mounting base at 25 °C.

455

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Nexperia | A Power Design Engineer's Guide

5.4 Continuous and pulsed currents, ID, IDM, IS
and ISMThe 25 °C current ratings are the same for both current directions (ID and IS).
D

ID

D

G

G
S

S

aaa-029304

aaa-029304

IS

The maximum current at any Tmb is the current which increases Tj to the maximum
allowed temperature (175 °C).

456

6 GaN FET static characteristics
These are the device parameters that explain how the GaN FET behaves in it normal
operating conditions

6.1 Gate-source threshold voltage, VGS(th)Gate-source threshold voltage for the GaN FET device is the gate-source thresholdvoltage of the Si MOSFET. This gate threshold is 3.9 V typically, with a negative
temperature co-efficient.
D

ID

Understanding Power GaN FET data sheet parameters

18

MOSFET & GaN FET Application Handbook

GS

ID - 1 mA; VGS = VGS(th)aaa-029304

Figure 9 | GaN FET gate-source threshold voltage

6.2 Drain-source on-state resistance, RDS(on)457

Understanding Power GaN FET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

I
RSNS

ID

DUT

V

VGS

VDS

VDS

VGS

2.5 µs.01 % duty cycle

aaa-029305

Figure 10 | Test circuit for dynamic RDS(on)		

t

480 V
VDD

t

aaa-029306

Figure 11 | Dynamic RDS(on) waveform

RDS(on) is the drain-to-source on-state resistance of the GaN FET. This has a dynamic
value for GaN FETs. When RDS(on) is measured immediately after turn-on, following
period of blocking high- voltage, this value will be slightly higher than normal. Thisis due to temporary charge trapping in the device structure. Nexperia have
optimised the device so that charge trapping and dynamic RDS(on) are minimised.
As can be seen in Figure 12, RDS(on) is shown as a normalised function of junction
temperature.
3
2.5
2
1.5
1
0.5
0

0

25

50

a=

75

100

125

R DS(on)R DS(on)(25 °C)

150 175
Tj(°C)

200

aaa-027810

Figure 12 | Normalised drain-source on-state resistance as a funtion of junction temperature458

7 GaN FET dynamic characteristics
These are the device parameters that explain how the GaN FET behaves in its
normal operating conditions

7.1 Gate charge QG(tot), QGS and QGD

Gate charge for GaN FET is defined in the same way as a normal MOSFET, because
the cascode arrangement of a GaN FET means that the gate is in fact the gate of a
LV MOSFET. Since the LV MOSFET is a relatively small device it has a small gatecharge suited to high speed switching.

7.2 Capacitances Ciss, Coss and Crss

18
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MOSFET & GaN FET Application Handbook

Traditional small signal capacitance is shown in Figure 13. The discontinuity (stepchange) in Figure 13 is where GaN HEMT pinches off. Capacitance relates charge tovoltage and energy to voltage. For a linear capacitance, the following fundamental
equations apply:			
Q = CV				CV 2
E= ––––––					2

459Understanding Power GaN FET data sheet parameters

18

Nexperia | A Power Design Engineer's Guide

104
C
(pF)
103
Ciss
102
Coss

10
Crss

1

1

10

102

VDS (V)103

VGS = 0 V; f = 1 MHz
aaa-028058

Figure 13 | Input, output and reverse transfer capacitance as a funtion of drain-source voltage; typicalvalues

The capacitances Co(er) and Co(tr) attempt to capture these relationships for anonlinear capacitance.

7.3 Effective output capacitance (energy
related), Co(er)The GaN FET capacitance parameter Co(er) is the effective output capacitance(energy-related) dependent on the drain voltage
Note that 0 ≤ VDS ≤ 400 V shown above, means as the voltage rises from 0 V to400 V.

460

30
EOSS
(µJ)
25
20
15
10
5
0

1/2 · 190 pF · V DS2
0

100

200

300

400

600
500
VDS (V)

aaa-029307

Figure 14 | EOSS as a function of VDS

At a specific VDS, a unique value of Coss satisfies the equation:e.g.

Understanding Power GaN FET data sheet parameters

18

MOSFET & GaN FET Application Handbook

By integrating COSS with respect to VDS, the result will be QOSS. If we then integrate
QOSS with respect to VDS, we will then arrive at EOSS.
EOSS * switching frequency will give the switching power loss in Watts.

7.4 Effective output capacitance (timerelated), Co(tr)The GaN FET capacitance parameter Co(tr) is the equivalent capacitance to givesame charging time, as VDS rises from 0 V to 400 V. This can also be described as aconstant current being used to charge the output capacitance, giving a time related
effective value.

461

Understanding Power GaN FET data sheet parameters

18

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Nexperia | A Power Design Engineer's Guide

At a specific VDS, a unique value of Coss satisfies the equation:e.g.

7.5 Output charge and stored energy, Qoss andEoss
Rather than using the Co(tr) and Co(er) parameters, it is much easier to use the graphs
shown below that are also available in the data sheet. The respective QOSS and EOSS
values can be read directly for the required VDS.
30
EOSS
(µJ)
25

160
QOSS
(nC)
120

20
15

80

10
40
5
0

0

100

200

300

400

500
600
VDS (V)700

aaa-029173

0

0

100

200

300

400

500
VDS (V)600

aaa-028063

Figure 15 | Typical QOSS			 Figure 16| Typical COSS stored energy

Understanding Power GaN FET data sheet parameters

18

MOSFET & GaN FET Application Handbook

7.6 GaN FET switching time characteristics

Nexperia GaN FETs generally require a ferrite bead in series with the gate. This
serves to de-”Q” the gate-source loop and improve the switching stability. The
ferrite bead effectively provides damping impedance at frequencies > 100 MHz. It
also introduces a small propagation delay, however it does not introduce any
additional loss.
The GAN063-650WSA in TO247 package has an integral ferrite bead, and so an
external bead is not required. Newer GaN FETs may not include the integral ferrite
bead, and so one must be included in the external gate drive. Always refer to thedata sheet of specific devices for details.

463

Understanding Power GaN FET data sheet parameters

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Nexperia | A Power Design Engineer's Guide

VDS

VGS

D

90 %

10 %

G
tR

tF

tD(on)tD(off)
tON

S

tOFF

aaa-029330

Figure 17 | Switching time waveform		

Figure 18 | Internal Ferrite bead

7.7 Source-drain voltage, VSD

This device parameter source-drain voltage refers to the GaN FET device when it isacting as a two terminal device when not enhanced with the gate VGS = 0 V.
150
IS
((A)
A)
125

D

Tj = 2
25°C
5°C

-

50°C
50°C
75°C
75°C
100°C
100°C

100

125°C
125°C
150°C
150°C

75

V SD

175°C
175°C
50

G

25
0

V GS = 0 V
0VGS = 0 V

2

4

6

8

VSD (V)

aaa-028059

Figure 19 | Source-drain (diode forward) current
as a function of source-drain (diode forward)
voltage; typical values

464

S

10

+

IS

Figure 20 | Source-drain (diode forward)
voltageVSD is the voltage developed whilst GaN FET conducts in reverse direction (e.g.
acting as a rectifier carrying freewheeling current) Here the voltage VSD comprises
the forward voltage of the Silicon MOSFET body diode and the voltage drop acrossthe 2 DEG channel of the GaN HEMT.

8 Qr for GaN FET switches

I, V
dlS/dt

IS

trr

IRM
DUT

A

Qr

0.25 IRMt

Understanding Power GaN FET data sheet parameters

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MOSFET & GaN FET Application Handbook

VRRM

VSD

+

aaa-029277

Figure 21 | Recovered charge test circuit and measurement waveforms

The Device Under Test (DUT) is carrying a freewheeling current. The switch in thehigh side of the test circuit in Figure 21 will then turn on and force a transition at
the switching node from low voltage to high voltage. This will cause the current in
the DUT to change from freewheeling current to some negative current up until
the point that the DUT is blocking the full DC supply voltage. If you integrate thenegative current you will get the minority carrier stored change plus any output
capacitance charge.
For GaN FETs, the charge on the output capacitance QOSS is the dominantcomponent of the reverse recovery charge Qr. Hence the reason that the datasheet
for the Nexperia GaN FET specifies the same value for both QOSS and Qr.
Some GaN device manufacturers will claim to have zero reverse recovery charge Qr
because they do not have a PN junction body diode. However, they will still have
QOSS.

465

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Nexperia | A Power Design Engineer's Guide

9 Switching-node snubber
To achieve maximum efficiency and stability when switching high currents, aswitching node RC snubber (R, Csn) is recommended. For IL < 14 A, a switching-nodesnubber is not required.
C sn 180
(pF)
160

V BUS

DC bus140
driver

120

RG

Q2

100
80

VS

60

driver

40

RG

Vo

Q1
RS N

RCSN

20
0

R C DCL(place as close as
possible to drain pin)

CSN

0

5

10

15

20

25

30
35
IL (A)

40

aaa-029331

R G = 30 Ω; τ = R SN × C SN = 1 ns aaa-029603

Figure 22 | Snubber capacitance as function of

Figure 23 | DC-link snubber circuit

load current

Note: a DC-link snubber is recommended in all cases. Optimal is 20 nF in series
with 4 Ω, most easily achieved with parallel combination 10 nF and 8 Ω. This
snubber lowers the Q factor of any resonance in the bus. That resonance will
act as a load on the high gain amplifier that is the GaN FET and can lead to
instability.
For some GaN FETs, like the GAN063-650WSA, an RC snubber is recommended
for the switching node. This will increase switching loss, so this is only
recommended at high power levels where the losses are a very small
percentage of the total power. Other GaN FETs do not require a snubber on theswitching node at any power level. Always refer to the data sheet of specificdevices for details.

How to read a GaN FET
datasheet voltage change466

10 Tables
Table 1.
Table 2.

Pinning information.......................................................................................450
Limiting values................................................................................................453

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468

Nexperia | A Power Design Engineer's Guide

10 Figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.

GaN FET Nomenclature................................................................................. 450
Conventional pin assignment....................................................................... 451
Nexperia pin assignment............................................................................... 451
Simplified GaN FET symbol........................................................................... 452
Duty Cycle Definition..................................................................................... 454
VTDS test circuit................................................................................................ 454
VTDS waveform................................................................................................. 454
Normalised total power dissipation as afunction of mounting base temperature................................................... 455
Figure 9. GaN FET gate-source threshold voltage..................................................... 457
Figure 10. Test circuit for dynamic RDS(on)...................................................................... 458
Figure 11. Dynamic RDS(on) waveform............................................................................. 458
Figure 12. Normalised drain-source on-state resistanceas a funtion of junction temperature.......................................................... 458
Figure 13. Input, output and reverse transfer capacitanceas a funtion of drain-source voltage; typical values.................................. 460
Figure 14. EOSS as a function of VDS................................................................................ 461
Figure 15. Typical QOSS...................................................................................................... 463
Figure 16. Typical COSS stored energy............................................................................ 463
Figure 17. Switching time waveform............................................................................. 464
Figure 18. Internal Ferrite bead...................................................................................... 464
Figure 19. Source-drain (diode forward) current as a function of source-drain(diode forward) voltage; typical values....................................................... 464
Figure 20. Source-drain (diode forward) voltage......................................................... 464
Figure 21. Recovered charge test circuit and measurement waveforms................ 465
Figure 22. Snubber capacitance as function of Load current. .................................. 466
Figure 23. DC-link snubber circuit................................................................................... 466

Chapter 19
An insight into Nexperia
Power GaN technology –Applications, quality,
reliability and scalability
Application Note: TN90004

An insight into Nexperia Power GaN technology – Applications, quality, reliability and scalability

19

MOSFET & GaN FET Application Handbook469

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Nexperia | A Power Design Engineer's Guide

1 Introduction
In this chapter we have concentrated on the Power GaN product and technology
robustness, quality, reliability and volume manufacturability based on GaN on Si
base material. We report here on the product robustness from applications andparametric point of view and qualification testing results of latest generation 650V
GaN FETs qualified in accordance with AEC-Q101 standard. Devices offer rated
operating temperature of -55°C to +175°C and this family comes with robust gatewith high threshold voltage which provides a high safety margin against
gate-source transients. Some of the tests are performed beyond AEC-Q101
requirements.
Most efficient power conversion requires best semiconductor devices as thefundamental building block. Power GaN technology offers best possible efficiency.
Proving performance is not in question for those closely following the Power GaN
technology for its performance demonstration in different applications. Providing
robustness of the products in operation, quality, reliability of the technology andthe scalability uptake in manufacturing are yet to prove. Power GaN technology
is full of challenges. In RF applications they are already successful but in highcurrent high power applications in high voltage, volume manufacturers need toaddress these challenges to satisfaction. In this paper we’d like to share productrobustness for applications, quality and reliability at high voltage and hightemperatures, for the Power GaN technology.
Nexperia Power GaN technology focuses on high power applications like AC/DC,
PFC, OBC, DC/ DC and Traction Inverters within 650V-900V for Automotive,
Telecom, Server, storage, data centres and industrial market sectors.
WBG materials with higher critical electric field and higher mobility together give
lowest RDS(on) (source drain on state resistance) for higher voltages and significantly
better switching FOM (Figure of Merit). WBG devices as beginning to enter themarket shows significant promise and takes away many limitations naturally
imposed by Si IGBT and Si SJ devices. Some of the hard switched application
topologies where Si SJ FETs cannot be used due to the diode reverse recovery caneasily use Power GaN FETs and take full advantage of reduced components countand higher efficiency with simpler control schemes.
GaN HEMT (High Electron Mobility Transistor) works with the formation of 2DEG (2Dimensional Electron Gas) due to the spontaneous polarisation and piezoelectric
polarisation combined at the interface of GaN and AlxGa1-xN. Epi is formed on Si
substrate via seed layer and a graded layer of GaN and AlGaN layers before thepure GaN layer is grown and thin layer of AlGaN then forms the 2DEG. Electron
mobility in this layer is very high hence the name.
Current Power GaN FETs are of two main flavour: E-mode or single die normally off

470

device, and D-mode or two die normally off device. Stability and leakage current of
the E-mode gates are of concern but two die normally off or cascode configuration
currently gives peace of mind as the driving of these FETs are simple and robust.
E-mode device drive is complex, especially for high voltage high power applications.
For high voltage and high power applications, to avoid gate bounce and harmful
shoot through situation, need to have high gate threshold voltage and stable gate
drive without worrying for over drive. This is currently not achievable with existing
E-mode technologies. The device in our presentation is two die normally off
configuration.

2 Product robustness
The product parameters shared here came from our 50 mΩ (typical at 25 °C) 650 V
device but all our products/technology share the same common robustness.
Product robustness include parametric assurance like, high reliability gate structure(± 20 V) and high threshold voltage (Vth = 4 V) that provides a high safety margin
against gate source transients induced due to the high drain source dv/dt, highvoltage source-drain transient specification (800 V for 650 V device) capable ofhandling switching transients up to 800 V reliably, rated operating range of -55 °C
to 175°C with Tj(max) of 175 °C, body diode characteristics with very low VF (1.3V @12 A) enabling Si- like freewheeling current capability without complex dead time
adjustments and along with other parametric performances are part of the productrobustness of the GaN product family.

3 Quality / qualification of power GaN
technology including failure modes
The latest generation 650 V GaN FETs are qualified in accordance with AEC-Q101
Rev D level qualification tests. Results shared here are done on 50 mΩ (typical at
25 °C) 650 V device that include 650 V, 175 °C High Temperature Reverse Bias
(HTRB) tests (1000 hrs) and dynamic RDS(on) shifts. In the following, the term
dynamic RDS(on) will be used to emphasize that RDS(on) measurements are made witha dynamic, switch-mode test. Temperature cycling tests (1000 cycles) are
performed over the range of -55°C to 150°C. High temperature (175 °C) Gate
positive (+20 V) and negative (-20 V) bias tests were performed. Further life tests
include high temperature biased and unbiased humidity tests and operating life
tests. These are only some of the critical tests performed and passed to show thereliability and high quality of the technology.

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Nexperia | A Power Design Engineer's Guide

3.1 HTRB
High-Temperature Reverse Bias has been performed at full rated voltage andmaximum operating temperature: 650 V and 175 °C. The condition for passing isthat RDS(on) does not shift by more than 20%. For GaN FETs it is important that
RDS(on) be tested dynamically to detect any short-term change due to charge
trapping. Figure 1 shows the shift in dynamic RDS(on) for the test population. Notethat the maximum shift is less than 15%. An additional HTRB test was performed
and passed at 800 V for 10 hours. This voltage is well above the DC rating, but does
correspond to the repetitive transient voltage rating.
40
R DS(on)(mΩ)

Measured values
Initialmeasurements
168 hours
504 hours
1008 hours

30
20

high limit
low limit

159

157

155

153

151

149

147

145

143

141

139

137

135

133

131

129

127

125

123

121

119

117

115

113

111

109

107

105

99

103

97

95

Control parts

101

93

91

89

87

85

83

81

259

257

255

0

253

10

251

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19

Parts on test

Figure 1 | Dynamic RDS(on) measurement during HTRB

3.2 HTOL
High-Temperature Operating Life tests are not part of the AEC-Q101 standard, but
are useful in validating reliability of the parts under actual operating conditions.
This is particularly important for new materials, like GaN, to ensure that any new or
unfamiliar failure modes are uncovered. A basic half-bridge operating in continuous
conduction mode provides the most fundamental exercise of switching behaviour.
For this test, a number of identical half-bridge circuits were prepared using two
each of the GAN063-650WSA. These were operated continuously as synchronousboost converters with the following conditions:
•
•
•
•
•

Vin = 200 V
Vout = 480 V
Pout = 800 W
Tj = 175 °C
Frequency = 300 kHz

The following graph shows efficiency of all samples during the 1,000 hour test. As
may be seen, there is no indication of degradation in any of the sample circuits.
Following the tests, all devices were tested for shifts in dynamic RDS(on), leakage
current, and threshold voltage. All parameters were found to be stable, with any

472

parametric shift within allowed levels.
100
Efficiency
(%)
99
98
97
96
95
0

100

200

300

400

500

600

700

800

900

1000

1100

time (hours)

Figure 2 | Efficiency of boost converters (HTOL) over 1000 hours

P

loss

(W)

9

Reference device (Si)8GaN FETs

7

6

0

750

1500

2250

3000
Time (hours)

Tested in hard-switched boost converter.
Operating conditions: 200 / 400 V converter operation:
175 °C; 300 kHz; 410 W
Figure 3 | Hard switched boost converter (HTOL) @ 175 °C for 3000 hours

Failure modes addressed by life testing power GaN technology failure modes canbe linked back to the life testing and extra testing performed. Like power cycling
relates to wire bond heel crack, temperature cycling address the solder
degradation, HTGB look at the gate insulator failure, HTRB addresses the field plate
insulator failure while high temperature direct current test addresses the trappedcharge in gate region and consequently higher RDS(on) (>20% assumes fail).

An insight into Nexperia Power GaN technology – Applications, quality, reliability and scalability

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MOSFET & GaN FET Application Handbook473

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19

474

Nexperia | A Power Design Engineer's Guide

4 Conclusion
Power GaN technology, though at the early stage of technology maturity offers
significant commercial viability with robustness and reliability. For scaling up,
tremendous growth potential is better addressed with power GaN technology on
Si.

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Power GaN the need for
efficient power
conversion

GaN FETs delivering
AEC-Q101 levelrobustness

CCPAK copper clipcomes to high voltage
applications

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Power GaN FETs astrategic approach tobring the best withoutcompromisingrobustness and quality

Nexperia copper-clip
SMD CCPAK GaN FET
package in half-bridge
evaluation board

5 References
1. 600 V JEDEC-qualified highly reliable GaN HEMTs on Si substrates- T. Kikkawa et al., 2014 IEEE International Electron Devices Meeting,
San Francisco, CA, 2014, pp. 2.6.1-2.6.4.
2. 650 V Highly Reliable GaN HEMTs on Si substrates over multiple 			generations: matching Silicon CMOS manufacturing metrics and process 		
control- S. Chowdhury, et al., Compound Semiconductor Integrated Circuit Symposium(CSICS), IEEE, 2016.

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Nexperia | A Power Design Engineer's Guide

6 Figures
Figure 1. Dynamic RDS(on) measurement during HTRB.............................................. 472
Figure 2. Efficiency of boost converters (HTOL) over 1000 hours.......................... 473
Figure 3. Hard switched boost converter (HTOL) @ 175 °C for 3000 hours......... 473

Chapter 20
Power GaN technology: theneed for efficient power
conversion
Application Note: AN90021

Power GaN technology: the need for efficient power conversion

20

MOSFET & GaN FET Application Handbook

477

Power GaN technology: the need for efficient power conversion

20

Nexperia | A Power Design Engineer's Guide

1 Introduction
Power Gallium Nitride(GaN) technology shows the greatest performance benefits
against other incumbent technologies including silicon (Si) technology. This chapter
details the scalability and growth of the technology as well as the quality andreliability of Nexperia GaN FETs. The performance of GaN FET power switches invarious applications is examined.
The biggest motivation and driver for semiconductor power device innovation is
improved efficiency in power conversion. Power Gallium Nitride (GaN) technology
shows the greatest performance benefits against other incumbent technologies,
including silicon (Si) technology. A reduction in power losses is the key challenge
industries are facing. Pressure from society and government legislation for reduced
CO2 emission are all trending towards more efficient power conversion andelectrification.
Automotive electrification, telecom infrastructure, server storage and industrial
automation using power electronics are the biggest trends in the technology
sector. GaN field-effect transistors (FETs) can enable us to achieve the best
efficiency with lower system costs while making the system lighter, smaller andcooler. Specifically, electrification in the automotive sector can be the biggest
beneficiary of this new power GaN FET technology.
To make all-electric vehicles (xEVs) a real replacement for vehicles that havefossil-fuelled internal combustion engines, efficient power conversion is key. There
are many challenges, such as range anxiety, harmful emissions, power losses, heatdissipation, cooling systems, system weight and power densities. On this journey,
industry is working in many ways to address these major issues to reduce range
anxiety and harmful emissions. These are integral parts to make the xEV platforms
a commercial and social reality.
Application areas in which there is a drive for significant improvement include:• Improved more efficient power conversion• AC-to-DC onboard charging• DC-to-DC power conversion• DC-to-AC inverter to drive the traction motors• Improved power density• Simpler driver and control scheme• Improved traction motors• Improved efficiency• Better torque and power• Lower losses• Higher dV/dt handling• Improved batteries, storage and battery management systems

478

Efficient power conversion helps significantly in many ways. An example is a
200 kW inverter with 95% efficiency versus 99% efficiency. A reduction in power
loss in full load is from 10 to 2 kW, about one-fifth, which means a significant
decrease of the cooling system. Not only that the loss is 8 kW lower (which can beeffective for useful traction power), but the smaller cooling also can help with
reducing the cooling energy consumption and on size and weight. Hence, a longer
range or smaller battery can be achieved.
This highlights the importance of achieving very high efficiencies. Power GaN FETs
based on GaN on Si epitaxy (epi) help significantly in this area, not only by offering
higher efficiency but also the most desirable scalability for growth to support thexEV growth ambitions. The simple Si fabrication (fab) process steps also allow thebest cost roadmap for commercial viability. Power GaN technology currently serveswith 650 V products for 400 V battery voltage and can serve up to 800 V battery
systems with up to 1,200 V power GaN devices. The power range can be up to 300
kW.

2 Power GaN FET switches
High-voltage (HV) power semiconductor switches are the fundamental building
blocks of any power conversion. Si-based insulated-gate bipolar transistors (IGBTs)
are currently dominating this market with significant maturity in the absence of any
better alternatives. The improvement of Si IGBTs and combining them with siliconcarbide (SiC) diodes helps to achieve incremental higher efficiencies, but has a limit
on how much further improvement is possible. Si IGBTs are fundamentally limited
in frequency of operations, speed and poor high-temperature performance alongwith poor low- current characteristics. Si super-junction (SJ) technology isdominating the power conversion at higher frequencies like AC-to-DC power factor
correction (PFC) and DC-to-DC power conversion. To achieve higher efficiency,
these devices are fundamentally limited to size and cost due to their inherent
material limitation for high-frequency operation. These limitations can besummarised by switching crossover losses, conduction, and reverse recovery losses.

20Power GaN technology: the need for efficient power conversion

MOSFET & GaN FET Application Handbook

In contrast however, wide band-gap (WBG) materials like GaN and SiC, assummarized in Table 1 are free from reverse recovery loss and can offer very low
switching cross-over losses (due to very fast turn on and off characteristics) andlower conduction losses. WBG materials with a higher critical electric field andhigher mobility together give the lowest drain source on-state resistance (RDS(on))
for higher voltages and a significantly better switching figure of merit. The WBG
devices beginning to enter the market show significant promise and remove manyof the limitations naturally imposed by Si IGBT and Si SJ devices. Some of the
difficult switched-application topologies where Si SJ FETs cannot be used due todiode reverse recovery can easily use power GaN FETs and take full advantage ofthe reduced component counts and higher efficiency with simpler control schemes.

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Power GaN technology: the need for efficient power conversion

20

Nexperia | A Power Design Engineer's Guide

The faster switching speeds and higher frequencies of operation enabled by GaN
power transistors help improve signal control, higher cutoff frequencies for passive
filter designs, and lower ripple currents, allowing for smaller inductors, capacitors,
and transformers. Consequently, the compact and smaller system solution offers
cost savings.

Table 1. Material properties

Material

Band Gap(eV)

CriticalElectronElectric Field mobility µ0
Ec (MV/cm)(cm2/Vs)

ThermalSaturation
conductivity velocity(W/cmK)(cm/s)

GaN

3.4

3.5

2,000

1.3

2.5 x 107

SiC

3.3

2.2

950

3.7

2 x 107

Si

1.12

0.23

1,400

1.5

1 x 107

There are two main options for current power GaN FETs: enhancement mode
(E-mode) or single die normally off devices and depletion mode (D-mode) or
two-die normally off devices. The stability and leakage currents of the E-mode
gates are of concern but the two-die normally off or cascade configuration
currently offers peace of mind as driving these FETs is simple and robust. The
E-mode device drive is complex, especially for HV, high-power applications. For
these applications, to avoid gate bounce and harmful shoot-through situations, it isnecessary to have a high gate threshold voltage and stable gate drive withoutworrying about overdrive,which currently is not achievable with existing E-mode
technologies. For operations up to 1 MHz switching frequency, cascode GaN FETs
are best suited, although current HV power-conversion frequencies are around only
300 kHz and traction inverter frequencies are still below 40 kHz. The GaN on Si
two-die normally off configuration allows significant design flexibility. Nexperia
GaN FET offers a ±20 V gate rating with an oxide/insulator gate, 4 V gate thresholdvoltage with 0 V turn off, and low gate charge. Hence, simple Si drivers are suitablefor use with these devices and for 0 – 8, 10, or 12 V, any gate drive can be used. In
contrast, the SiC technology generally requires at least 15 V, a very-high-current
driver with a negative gate drive capability to turn off the device adds costs for thedriver and increased driver and switching losses. Nexperia GaN also brings a very
good anti-parallel diode built-in that helps with the robust freewheeling conduction
path. The cascode version offers significant freedom to make the gate structure
have the same robustness that automotive customers are used to. This is valid forboth Si FET and the GaN high electron mobility transistor (HEMT) with insulated
gates.
Figure 1 shows the cross-section for the GaN HEMT, which works with theformation of the 2D electron gas (2DEG) due to the spontaneous and piezoelectric
polarization combined at the interface of GaN and AlxGa1-xN. GaN Epi is formed on480the Si substrate via the seed layer and a graded layer of GaN and AlGaN layers
before the pure GaN layer grows. A thin layer of AlGaN then forms the 2DEG.
Electron mobility in this layer is very high, hence the name.
Gate Dielectric
SourceD

Field plate/s
Gate

Drain
Dielectric

AlGaN
G

GaN

2DEG Channel

Si SubstrateS

aaa-028116

aaa-032042

Figure 1 | GaN HEMT cross section
			

Figure 2 | GaN HEMT graphic
symbolFigure 2 shows how combining a low-voltage (LV; 30 V) robust Si MOSFET with thecascode configuration can eliminate all of the major concerns regarding the poorgate structures of the E- mode devices and make the entire usage very simple. An
LV Si-based gate structure is a very mature technology and engineers areaccustomed to using them.
The reverse recovery charge (Qrr) for the cascode power GaN is very low as shownin Figure 3 and mostly capacitive and allows the full potential of the power GaN
technology. Actual reverse recovery charge (Qrr) for the LV Si FET used here is only
~12 nC. The E-mode device channel mobility is much lower compared to thechannel mobility in a D-mode GaN HEMT channel.

GaN FETR everse turn-off, 41 mΩ
400 V, 40 A, 800 A/μs, Qrr = 179 nC

Power GaN technology: the need for efficient power conversion

20

MOSFET & GaN FET Application Handbook

Si MOSFET
R everse turn-off, 41 mΩ
400 V, 40 A, 800 A/μs, Qrr = 2790 nC

Figure 3 | Qrr, reverse recovery charges for GaN FET vs Si MOSFET

481

Power GaN technology: the need for efficient power conversion

20

Nexperia | A Power Design Engineer's Guide

Driving a cascode device is also very simple. Cascode device operation is shown in
Figure 4 and Figure 5 for different bias situations. Power GaN FETs can be used in abi-directional form and also allow simpler bi-directional power conversion.

D

D

D

G

G

G
S

S

S

Forward conduction

Reverse conduction 1(VGS < Vth)Reverse conduction 2(VGS > Vth)aaa-032059

Figure 4 | GaN FET operation

I
ID
(A)

200

I

150

VGS = 10 V
8V
7V

100

50

6V
5V

0

-50

III
VGS = 0 V

-100

VGS = 8 V

-150
-10

-8

-6

-4

-2

0

2

4

6

8
10
VDS (V)

aaa-032043

Figure 5 | Cascode GaN HEMFET forward and reverse output characteristics

Since power GaN transistors were introduced to the market, significant
improvements in performance, reliability, cost, and availability have taken place.
More capable GaN power transistors are becoming available to drive higher power
to be compatible with the EV requirements while being well positioned for
applications in data centres, telecom infrastructures and industrial.

482

3 Scalability and growth
Si wafers are widely available in different sizes. GaN on Si wafers allow us to use
150 mm, 200 mm metal organic chemical vapor deposition (MOCVD) reactors for
epi growth and can be processed in mature silicon semiconductor fabs. There arelots of Silicon fab capacity that can be utilised. Processing is similar to standard Si
processing, which is not so complex. Tremendous growth potential is better
addressed with power GaN technology on Si at a right cost point.

4 Applications and performance
Whether it is AC-to-DC PFC stage, a DC-to-DC converter Figure 6 or traction inverter
Figure 7 the basic building block for most topologies is a half-bridge (Figure 9).
Hence when GaN FETs are compared against Si FETs in a simple boost converter,
GaN FET shows its superior performance due to the differences in material
properties (Figure 8). All of these applications can take advantage of these benefits
and hence reduce losses. The advantages of power GAN FETs compared to Si IGBTs
come from the light load, high temperatures and higher frequencies when thelosses are very high.

Isolated DC-DC(Bidirectional)

PFC
400 VDC BusPower 6.6 kW

GaN
FET

Si
FET

GaN
FET

GaN
FET

GaN
FET

GaN
FET
HV
Battery(280 VDC
To
420 VDC)

240 VAC
Single
Phase

GaN
FET

Si
FET

GaN
FET

GaN
FET

AC-DC Converter for OBC
and DC-AC Inverter

GaN
FET

Power GaN technology: the need for efficient power conversion

20

MOSFET & GaN FET Application Handbook

GaN
FET

aaa-029378

Figure 6 | AC-to-DC PFC stage and isolated DC-DC configuration

483

Power GaN technology: the need for efficient power conversion

20

Nexperia | A Power Design Engineer's Guide

VDC Bus
GaN
FET

GaN
FET

GaN
FET

s
GaN
FET

GaN
FET

GaN
FET

CABLE
PSMN

TRACTION INVERTER

aaa-032041

Figure 7 | Traction inverter

Q2
L1

Vin
C1

VS

VO
C2

Q1

GND

Vin = 240 V, Vout = 400 V, fsw = 100 kHz
Same converter with replacement
switches
GaN
Si

Figure 8 | Half-bridge boost converter(GaN switch vs Si CoolMOS)

aaa-032060

Figure 9 | Half-bridge boost converter

GaN FET losses are significantly lower due to the absence of reverse recovery losses
and switching crossover losses. It is possible to achieve near ideal turn on and off
losses with dV/dt of approximately 200 V/ns.
GaN switches are extremely fast and routinely used in radio-frequency amplifiers at
gigahertz frequencies, although at much lower voltages but show the capability as
both use the GaN HEMT structure. Since the GaN devices are very fast and can beused in applications with high dV/dt and high dI/dt, care must be taken to optimize
the PCB layouts. To minimize parasitic inductances layout optimization is absolutely
fundamental to Power GaN usage. To maximize the performance of surface-mount484

packages and low inductance, high-current, high-performance modules are
essential and being worked on. Currently, using GaN devices in a traction invertermeans slowing it down significantly to save the motor windings. They are nearly
limited to a dV/dt of 10 V/ns, which holds significant potential to improve themotors and take the frequency up to 40 kHz to improve the power density
significantly. The development of new electric motor technology with improved
ability is on the way to make the system more efficient.

5 Quality and reliability
Power GaN FET technology currently showing good quality and reliability as
multiple vendors demonstrated Joint Electron Device Engineering Council (JEDEC)
and Automotive Electronics Council (AEC) Q101 quality standards. To demonstrate
the reliability of power GaN technology, these are minimums and must be fulfilled.
For GaN, only the existing quality standards are not enough as the material is new
and operates differently. Dynamic RDS(on) or current collapse phenomena is wellknown for power GaN FETs and this method for testing is introduced and devices
are verified. Material quality, trapping and appropriate de-trapping that is
responsible for the dynamic RDS(on) can be measured and given high-level
confidence for its usage as the values getting better and are now around 10%. We
need to continuously look at the different failure modes and take the devices tofailure and understand the physics of failure.
Beyond AEC-Q101 qualifications, for validating the GaN FETs’ reliability in actual
operating conditions, several identical half-bridge circuits (with continuous current
conduction mode) were prepared using one high and one low side GAN063650WSA. These were operated continuously for 1000 hours as synchronous-boost
converters with the conditions: Vin = 200 V, Vout = 480 V, Pout = 800 W, Tj = 175 °C
and frequency = 300 KHz. There is no indication of any degradation in theperformance of any circuits for all samples for the whole duration of 1000 hours of
test.

20
Power GaN technology: the need for efficient power conversion

MOSFET & GaN FET Application Handbook

Following the high temperature switching tests, all devices were tested for shifts indynamic RDS(on), leakage current, and threshold voltage. All parameters were foundto be stable, with any parametric shifts below the allowed levels.
The device specification has 800 V transient voltage capability guaranteed by test
to eliminate any concerns for over voltage spikes Similarly many other over stresses
like voltage and temperatures are used and different acceleration factors aredefined for end of life and Failure in Time (FIT) rates estimated for the application
situations. As the volume of products shipped increases, real field failure rates will
be determined.
Power GaN technology is ready to take its place for efficient power conversion. We
are beginning to see its adoption in the non-auto segments and soon it will be used

485

Power GaN technology: the need for efficient power conversion

20

486

Nexperia | A Power Design Engineer's Guide

in automotive applications which can take advantage of lower losses and higher
power densities. Si technology is well established in the market but reaching its
limit. Power GaN technology will be the norm in the future as fear of unknown isreduced going forward.

6 References
1. AN90005 - Understanding Power GaN FET data sheet parameters
2. AN90006 - Circuit design and PCB layout recommendations for GaN FET
half-bridges

7 Tables
Table 1.

Material properties......................................................................................... 480

20Power GaN technology: the need for efficient power conversion

MOSFET & GaN FET Application Handbook

487

Power GaN technology: the need for efficient power conversion

20

488

Nexperia | A Power Design Engineer's Guide

8 Figures
Figure 1. GaN HEMT cross section................................................................................ 481
Figure 2. GaN HEMT graphic symbol............................................................................ 481
Figure 3. Qrr, reverse recovery charges for GaN FETvs Si MOSFET................................................................................................... 481
Figure 4. GaN FET operation.......................................................................................... 482
Figure 5. Cascode GaN HEMFET forward and reverse
output characteristics.................................................................................... 482
Figure 6. AC-to-DC PFC stage and isolated DC-DC configuration........................... 483
Figure 7. Traction inverter............................................................................................. 484
Figure 8. Half-bridge boost converter(GaN switch vs Si CoolMOS).......................................................................... 484
Figure 9. Half-bridge boost converter......................................................................... 484

Chapter 21
Circuit design
and PCB layout
recommendations
for GaN FET
half bridges
Application Note: AN90006

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

489

Circuit design and PCB layout recommendations for GaN FET half bridges

21

490

Nexperia | A Power Design Engineer's Guide

1 Introduction - Diode-free bridges
This chapter explains the recommendations for circuit design and PCB layout when
applying GaN FET half bridges.
Power GaN FETs are nearly ideal switches for many applications. A particularadvantage in bridge circuits is that they can carry the freewheeling current without
the need of an additional anti-parallel diode. The diagrams in Figure 1 compare a
traditional high-voltage half bridge to a half bridge made with GaN FET devices. In
the traditional half bridge each switch (shown here as an IGBT) is paired with afreewheeling diode. Because the HEMT channel exists in pure, undoped GaN, there
is no parasitic p-n junction to provide an unwanted current path, and bidirectional
flow of majority carriers can be realized in the channel.
Vdd

Hi/Lo
DRIVER

Vdd

Hi/Lo
DRIVER

aaa-029628

Figure 1 | Comparison of a traditional high-voltage half bridge to a GaN FET half bridge.

In Nexperia’s two-die GaN FET the freewheeling current does indeed flow in the
body diode of a silicon MOSFET, but because the silicon MOSFET is a low voltage
part, the injected charge is very small. Indicated in Figure 2 are the current paths
for three modes of operation. In the reverse conducting mode the conduction loss
may be reduced by enhancing the silicon MOSFET (driving VGS > VGS(th)). As indicatedin Figure 2, the voltage drop from source to drain decreases by about 0.8 V with a
5 A reverse current when the gate is enhanced.

aaa-029630

aaa-029631

b. Reverse conduction
VGS < VGS(th)aaa-029629

a. Forward conduction

c. Reverse conduction
VGS > VGS(th)

15
(A)

(1)

10

(2)

5

(3)

I

0

III

(4)

-5
(5)

-10
-15

-3

(1) VGS = 8 V(2) VGS = 4 V

-2

-1

0

1

(3) VGS = 3 V(4) VGS = 0 V, 1 V, 2V(5) VDS

2

(VDS)

3

aaa-029632

Figure 2 | Current paths in the GaN FET switch for three operating modes, and the corresponding I-V
characteristic.

Some transistor technologies include junctions which could, if permitted, serve thefunction of the freewheeling diode, the body diode of a MOSFET being one
example. The reverse recovery charge for these devices will be much larger thanfor a GaN FET with similar ratings. The graphs shown in Figure 3 compare the total
reverse-recovery charge for a Nexperia GaN FET to a low-Qrr (CFD series)
superjunction silicon MOSFET. The Qrr ratio, Si/GaN, varies with exact part numbers.

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

491

Circuit design and PCB layout recommendations for GaN FET half bridges

21

492

Nexperia | A Power Design Engineer's Guide

(V)

60

600
Vd (V)Id (A)

500

50

(A)

(V)

60

600
Vd (V)Id (A)

500

50

400

40

400

40

300

30

300

30

200

20

200

20

100

10

100

10

0

0

0

0

179 nC

-10

-100

-200

-20

-200

-300

-30

-300

-400

-40

-400

-50
300

-500

-100

-500

0

50

100

150

200

250

(ns)

a. GaN FET

(A)

-10
-20

2790 nC

-30
-40

0

50

100

150

200

250

(ns)

-50
300

b. CFD series superjunction Si MOSFET
aaa-029633

aaa-029634

Figure 3 | Reverse recovery charge test result for a GaN FET (left) and a CFD series superjunction Si
MOSFET (right).

2 Circuit-design recommendations
The high switching speed of GaN FET devices necessitates observation of a few
specific circuit-design guidelines. Before explaining these requirements, however,
one simplification may be mentioned: negative gate drive is neither necessary nor
recommended. Due to the high threshold voltage (Vth = 4 V typical) and extremely
low Miller capacitance (QGD = 6 nC typical), there is adequate turn-off margin with asimple 0 V to 10-12 V drive.
While high-speed switching brings the benefit of reduced power loss, the inherent
transients can create stability problems. Specifically, the high transient duringswitching, combined with parasitic inductances, leads to transient voltages in thecircuit. These voltage transients can interfere with the gate and the driver of thedevice, and, in the worst case, creates sustained oscillation that must be prevented
for safe operation of the circuit. The following section provides guidance on how toeliminate oscillation and how to achieve high switching current with acontrolled .

VBUS

DC busdriver

RG

Q2

RCDCL
(place as close as
possible to drain pin)VSdriver

RG

Vo

Q1
RSN

RCSN

C SN

aaa-029331

Figure 4 | Half-bridge switching circuit

2.1 Solutions to suppress oscillation
To avoid sustained oscillation, it is important to minimize noise generation, tominimize noise feedback, and to damp the ringing energy resulting from the highcurrent and voltage transients. This can be achieved with the recommendations
outlined below using a half-bridge switching circuit in Figure 4 as an example.
1. Optimize the PCB layout to minimize external parasitic inductances and 		associated feedback. Details follow in Section 3 of this chapter.
2. Use a DC-link RC snubber [RCDCL in Figure 4]. The DC rail or DC-link, when
decoupled with a low-ESR fast capacitor, can be considered a high-Q C-L 		
network at high frequencies (with "L" being the feed inductance of the DC bus).
This can interact with the devices during transients and lead to ringing. Adding
an RC snubber across the DC-link close to the drain pin of the high- side device
can effectively absorb the ringing energy, suppressing potential oscillation. Thiseffect can be seen where the high-frequency ringing at 25 A turn-off is 		
substantially damped with the RCDCL. Since this snubber is not inserted at theswitching node, it does not add switching loss to the circuit.

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

Note: This is recommended even for single-ended non-half-bridge 		
designs. The practical values of the RCDCLcan be two sets of 6 - 10 Ω / 0.5 W
SMD resistors in series with a 10 nF / 600 - 1000 V ceramic SMD capacitor, or
one 3 - 4 Ω / 1 W resistors in series with a 10 - 20 nF / 600 - 1000 V capacitor
if space is limited.
3. Adding a switching-node RC snubber [RCSN in Figure 4] can further reducehigh-frequency ringing and help control
transients at high operating 		

493

Circuit design and PCB layout recommendations for GaN FET half bridges

21

Nexperia | A Power Design Engineer's Guide

currents. The effect of the RCSN on the switching waveform can be seen at aswitching current > ISWL, see Table 1. Unlike the RCDCL, the capacitance of the
RCSN does increase switching loss. The degradation in efficiency is minimal
however, when using the recommended snubber parameters given in the data
sheet, and summarized in Table 1.

2.2 Gate drive and bootstrap supply
Since high slew rates of the order of 50 - 100 V/ns are normal, the high-sidegate-driver must have good common-mode transient immunity. Apart from thatconsideration, there are no special requirements for the Hi/Lo gate driver used with
Nexperia’s GaN FET switches. As with any insulated-gate power transistor, thegate-drive current should be consistent with the desired turn-on time and total
gate charge. As mentioned in the opening paragraph of Section 2, use of a negativegate voltage in the off state is not recommended. Selecting a gate driver with alower drive current can be appropriate for reducing . Drivers with 0.5 A output
current have been used with good results, for example. Nexperia GaN FETs in the
TO-247 package include an integrated ferrite bead in series with the gate.
The bootstrap, or floating, supply for the high-side gate drive comprises
components R4, D1, C12, and C13 in Figure 8. The junction capacitance of D1
contributes directly to switching loss, and so a fast, low capacitance diode should
be used. Resistor R4 is critical for limiting the inrush charging current; a value of 10- 15 Ω works well. If an isolated DC-DC converter is used for the high-side supply,
the isolation capacitance plays the same role as the junction capacitance of D1 inthe bootstrap supply. Inductance in series with this capacitance will create an
additional resonance which will be excited with each switching transient, so carefullayout applies here. Use of a common-mode choke in the floating supply can be
helpful.

2.3 Summary of circuit-design
recommendations• SMD mounting is recommended for all snubber components.
• A gate resistor (RG) is required for all devices.
• The RCDCL snubber reduces voltage ringing due to interaction of the GaN FET
with the bypass network.
• The RCSN snubber enables increased output power while slightly reducing lightand medium load efficiency.
• The RCSN implementation in a half-bridge has the advantage of allowing a higher
peak turn-off switching current due to the reduction of the seen by the 		
freewheeling device as the main active switch turns off.

494

• Gate ferrite beads (FB1) may be required for future devices in other package
types; TO-247 devices have built-in ferrite beads hence no external FB1 is 		needed.
• Refer to Nexperia Application Note AN90005 (chapter 18 in this book) - 		Understanding Power GaN FET data sheet parameters for an explanation ofdata-sheet limitations and relevant recommendations for when to use a 		switching-node snubber.

2.4 Required and recommended external
components
The recommended components of the half-bridge circuit in Figure 4 aresummarized in Table 1.
They have been tested and verified to prevent oscillation for safe, reliable
operation with the recommended gate drive voltage ranges shown. Using a higher“on” voltage is not recommended and may increase the propensity for oscillation
and will require a larger gate resistor. Using a lower “on” voltage may increaseswitching and conduction losses due to increased RDS(on).

Table 1. Recommended components for half-bridge circuit
Parameters \ Part Number
Package
Recommended Gate-Drive Voltage
Recommended Gate Resistor (Rg)
Gate Ferrite Bead (FB1)
Recommended RCSN
Recommended RCDCL
ISWL

GaN041-650WSA
TO-2470 V, 10 - 12 V
30 Ω
Not required
200 pF + 5 Ω(10 nF + 8 Ω) x 2
17 A

GaN063-650WSA
TO-2470 V, 10 - 12 V
30 - 45 Ω
Not required
100 pF + 10 Ω
(10 nF + 8 Ω) x 2
14 A

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

2.5 To verify GaN FET stable operation
To verify adequate operational margin without oscillation, as a minimum observe
the VDS waveforms at the turn-on and turn-off switching edges at the application’s
maximum drain current. This may occur during start-up or at the application’s
maximum load step. A double-pulse or multi-pulse test is highly recommended
utilizing the actual layout, with current levels at or greater than 120 % of theapplication’s anticipated peak current. Verify that the ringing on the VDS waveformat the transition edges is adequately damped.

495

Circuit design and PCB layout recommendations for GaN FET half bridges

21

496

Nexperia | A Power Design Engineer's Guide

3 PCB layout
The parasitic inductances of the input (gate) and output (power) loops contributesignificantly to overshoot, ringing, and stability in general.

3.1 Pin assignment
Nexperia GaN FETs in the TO-247 package use a Gate-Source-Drain pin assignment,
which differs from that used for older transistors in the same package. The reason
for centering the source rather than the drain is that the source is the common
node to both gate and power loops. This minimizes coupling of the two loops, as
illustrated in Figure 5.

G

gate
loop

D

S

G

ZL

power
loop

gate
loop

a. Conventional pins assginment

S

D

ZL

power
loop

b. Nexperia pin assigment

aaa-029293

aaa-029294

Figure 5 | Illustration of the de-coupling achieved with a centered-source pin assignment

3.2 Power loop
Although the various charges are low with GaN FET switches, they are not zero, andthere will be a very fast transient current during switching as these charges are
redistributed. This current will flow from the positive supply node, through bothtransistors to the negative supply node. To minimize ringing due to this transient,inductances in this path should be minimized. Referring to Figure 6, these
inductances are indicated as LS1, LD1, LS2, and LD2. To minimize these, low
impedance power and ground traces, or planes, should be used and bypass
capacitance and DC-link snubber should be placed as close as possible to thetransistors. It is not critical to minimize inductance Lout since it is in series with theload inductance, and to a first approximation simply adds to it. The connection
between the high-side (Q1) source pin and the low-side (Q2) drain pin should also
be very short. Placing the two transistors back-to-back on a common heat sinkhelps accomplish this. Shown in Figure 7 to Figure 10, is a portion of the layout of a
Nexperia half-bridge evaluation board, indicating the placement of the powertransistors

Vdd
LD1

LD_int1

LG1

LG_int1

LS_int1

+HVdc

LS1

outloop-looptransient current

LD1

LD2

Q1

Hi/Lo
GATE
DRIVERLout
Cout

L_loadL

LD_int2

LS1
Lout

Lload

LG2
LD2

LG_int2

Cout

LS_int2

Q2
LS2

LS2

a. Only elements external to GaN FET packages

aaa-029639

b. Both internal and external parasitic elements

aaa-029640

Figure 6 | Parasitic elements in the power loop

Details of the power loop layout are highlighted in Figure 9. To minimize
inductance, the drain of Q1 is connected directly to the power plane; the source of
Q2 is connected directly to the ground plane.
The power and ground planes are on internal layers of the PCB, not visible in the
figure. The switching node is formed by the wide trace connecting the source of Q1
to the drain of Q2.

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

497

Circuit design and PCB layout recommendations for GaN FET half bridges

21

Nexperia | A Power Design Engineer's Guide

Figure 7 | PCB layout - Power-stage
HV

R17
10 Ω

R18
10 Ω

R19
10 Ω

C33
10 nF

C34
10 nF

C35
10 nF

C4
10 nF

C5
10 nF

C6
10 nF

C7
0.1 µF

R3
10 MΩ

PGND

PGND

VG

+5 V

HV

R4

TP1

VIA

TP3

VIB
VDDI

PWM_U_LO

GND1
DISABLE
DT
n.c.

AGND

HV

D1

AGND

PWM_U_HI

TP4

10 Ω

C11
0.1 µFC10
0.1 µF

VDDI1

R7
12 kΩ

AGND

U2
1

16

2

15
14

3
4
5

VDDA
VOA

C12
0.1 µF

C13
10 µF

GNDA

Q1
GaN063-650WSA
R11
G
30 Ω

D
S

R13
15 Ω
C24
47 pF

R14
15 Ω

TP2
PFC103100-1

SI8230

6

11

7

10

8

9

VDDB
VOB

C14
0.1 µF

C15
10 µF

GNDB

AGND

Q2
GaN063-650WSA
R12
G
30 Ω

D
S

R15
15 Ω

R16
15 Ω
TP5

C25
47 pF

PGND

PGND

aaa-029641

Figure 8 | PCB layout - Corresponding schematic section

498

Figure 9 | PCB layout - Detail of power connections Figure 10 | PCB layout - Detail of bypassing:					

SMD capacitors on PCB backside

Illustrated in Figure 11 is an alternative placement where the GaN FETs are
side-by-side, rather than back-to-back, as in Figure 7 to Figure 10. Inductances arehigher with this arrangement, but it is sometimes preferable for manufacturing
reasons.
+5 V

TP1

VCC

C32

C33

C34

R22

1 µF

1 µF

10 µF

4.7 Ω

GND1

GND1

PWM_H

VIA

PWM_L
C78
10 pF

TP2

GND1
DISABLE

GND1

GND1

VIB
VDDI

C79
10 pF

+5 V
R69

R71

100 kΩ

0Ω

R24

DT

12 kΩ
(65 kΩ
for
8274)

VDDI1

n.c.

1

16

2

15

3

14

4
5

D3

AGND

U7

VDDA
R70

VOA
GNDA

GH

PB1

G

40
R45
10 kΩ

11

7

10

8

9

VCC

VDDB

R42

VOB
GNDB

C44
1 µF

C43
4.7 µF

GL

30 Ω

AGND

AGND

TP4
D

R28
15 Ω

DC OUT

S
C75
47 pF
R43
10 Ω

R44
10 Ω

R55
10 Ω
C48
10 µH

L6
HF-TOROID
C14
10 nF
Q3
GaN041-650WSA

GL

GH

30 Ω

VS

SI82306

GND1

Q2
GaN041-650WSA

C38
10 µF

C39
1 µF

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

FB2

G

40

TP8

R47
10 kΩ

D

R51
15 Ω

C22
10 nF

C23
10 nF

C62
680 µFC49
680 µFC72
680 µFC67
680 µF

TP5

S
C74
47 pF

5 4 3 2AGND(a) Schematic diagram

aaa-029642

499

Circuit design and PCB layout recommendations for GaN FET half bridges

21

500

Nexperia | A Power Design Engineer's Guide(b) PCB layout

Figure 11 | Totem pole PFC PCBCapacitance on the switching node, indicated as Cout in Figure 6, adds directly toswitching loss, and so the total area of copper which forms the switching node
should be kept low, but not so low that inductance becomes significant. A typical
4-layer board with 12 mils (0.3 mm) between outer trace and internal ground planeadds about 15 pF / cm2. At 100 kHz switching, each 1 cm2 would add
P = 1/2 CV2 fs = 120 mW switching loss, for example.

4 Heat Sink connection
The heat sink should be connected to an AC ground. In the evaluation board of
Figure 7 to Figure 10 the heat sink is electrically connected to the negative supply,or ground plane. Both low-side and high-side transistors are insulated from theheat sink. For the low-side transistor the capacitance between tab and heat sink is
not critical, since the tab of the TO-247 is connected to the source. This transistor
could be mounted directly to the heat sink, but the possibility that load current
could flow in the heat sink must be allowed for. If a reliable connection between
tab and heat sink is either not possible or not desirable, use of an insulator isnecessary. For the high-side transistor, capacitance between the TO-247 mounting
base and the heat sink will add to switching loss, and so a thick and/or low
permittivity insulator should be used.

4.1 Gate-drive loop
As with the output loop, minimizing inductance in the input, or gate-drive loop is
critical. Particularly important is the source inductance (LS2 in Figure 6, forexample), which is common to both loops.
Any voltage developed across this inductance due to a change in the output
current,, will appear in the input loop. The gate–source loop should be made ascompact as possible to minimise the gate loop inductance. The gate-source loop
inductance and GaN FET input capacitance form a resonant tank and will have acorresponding resonant frequency. The resonant circuit will be excited into
oscillation by the back EMF produced by the in the source inductance. Figure 12
Usually the resonant frequency of the gate-source loop will be less than 200 MHz. A
ferrite bead fitted in series with the gate can have a beneficial effect by “de-Qing”
the gate-source resonant circuit. For this reason, a small ferrite bead is integrated
inside the TO-247 package.
For future Nexperia GaN FETs where it is not integrated, the ferrite bead should belocated as close as possible to the gate pin of the GaN FET so that the damping isapplied to the device gate where it is needed. A gate ferrite bead is needed forboth single-ended non half-bridges as well as half-bridge designs.
Vdd

Ld

Ld_int

Cgd_GaN
Rg_int1

GaN
HEMT

Cds_GaN

M1
LV_MOS

Cds_MOSCgs_GaN

Circuit design and PCB layout recommendations for GaN FET half bridges

21

MOSFET & GaN FET Application Handbook

Mid-Node
Cgd_MOS

Vg

Rg

Lg

Lg_intRg_intCgs_MOSResonant circuit

Ls_intLs

aaa-029643

Figure 12 | Input loop for a cascode GaN FET

501

Circuit design and PCB layout recommendations for GaN FET half bridges

21

502

Nexperia | A Power Design Engineer's Guide

4.2 Summary of PCB layout recommendations• Place the RCDCL as close as possible to the drain pin of the high-side GaN FET
and ground it to the large ground plane.
• Use a large area ground plane for an overall low-noise base potential.
• Arrange the gate drive circuit on one side and the output circuit on the otherside of the PCB to minimize noise feedback from the output loop to the input
loop.
• Place the driver circuit close to the gate of the device.
• Shorten the power loop by arranging the high-side and low-side devices close-by.

Nexperia copper-clip
SMD CCPAK GaN FET
package in half-bridge
evaluation board

5 Tables
Table 1.

Recommended components for half-bridge circuit.................................495

21
Circuit design and PCB layout recommendations for GaN FET half bridges

MOSFET & GaN FET Application Handbook

503

Circuit design and PCB layout recommendations for GaN FET half bridges

21

504

Nexperia | A Power Design Engineer's Guide

6 Figures
Figure 1. Comparison of a traditional high-voltage half		bridge to a GaN FET half bridge.................................................................. 490
Figure 2. Current paths in the GaN FET switch for three operating modes,		and the corresponding I-V characteristic................................................... 491
Figure 3. Reverse recovery charge test result for a GaN FET (left)		and a CFD series superjunction Si MOSFET (right)................................... 492
Figure 4. Half-bridge switching circuit........................................................................ 493
Figure 5. Illustration of the de-coupling achieved with a centered-source		
pin assignment............................................................................................... 496
Figure 6. Parasitic elements in the power loop......................................................... 497
Figure 7. PCB layout - Power-stage............................................................................. 498
Figure 8. PCB layout - Corresponding schematic section........................................ 498
Figure 9. PCB layout - Detail of power connections................................................. 499
Figure 10. PCB layout - Detail of bypassing: SMD
		capacitors on PCB backside......................................................................... 499
Figure 11. Totem pole PFC PCB..................................................................................... 500
Figure 12. Input Loop for a cascode GaN FET.............................................................. 501

Chapter 22
Probing
considerations
for fast switching
applications
Application Note: AN90004

Probing considerations for fast switching applications

22

MOSFET & GaN FET Application Handbook

505

Probing considerations for fast switching applications

22

Nexperia | A Power Design Engineer's Guide

1 Introduction
Accurate voltage waveform measurements of fast switching circuits require special
care and attention in the use of the oscilloscope probe. This chapter details the useof a probe ground spring to achieve the best results.
The latest GaN FET devices from Nexperia are capable of very fast switching of highvoltages and currents. This requires special care to be taken in the measurement of
the switching waveforms.
In order to accurately measure voltages on fast switching nodes, best measurement
practices should be followed. It should be noted that unless efforts are made tofollow best practice as closely as possible, then the voltages seen on theoscilloscope may be an artefact of the measurement rather than a true
representation of the real voltage on the node.
An example of a fast switching waveform is shown in Figure 1 below. Note thedifference between measurements results with a) the oscilloscope probe groundconnection made using a standard probe ground wire and b) the samemeasurement with the ground connection made using a probe ground spring.
Ground spring

Ground wire

30
V
25
20
15
10
5
0
-5
-20

0

20

40

80
60
time (μs)

Figure 1 | Switching waveform; probe grounding method comparison

2 Oscilloscope probe ground spring
The ground lead length of the oscilloscope probe can have a very detrimental
effect on the measurements taken. If you use the standard 10 cm loop suppliedwith the scope probe, the measurements that you make will be prone to noise andpickup, as the loop acts as an aerial and picks up noise in proportion to the loop
area.

506

In order to make switching node measurements that are less prone to noise andpickup, the following type of oscilloscope probe connections should be made:Figure 2 | Probing a fast switching node

Note that the oscilloscope probe ground connection is made via a test probeground spring, see Figure 3. It is important that the ground wire is kept in parallelwith the probe tip and the loop area is kept to an absolute minimum.

22
Probing considerations for fast switching applications

MOSFET & GaN FET Application Handbook

Figure 3 | Probe ground spring

These oscilloscope probe accessories are available from electronic test equipment
distributors and retailers - e.g. Mouser Electronics Part No: 940-PK1-5MM-118 (Mfr.
Part No: PK1-5MM-118).

507

Probing considerations for fast switching applications

22

Nexperia | A Power Design Engineer's Guide

Figure 4 | Oscilloscope probe with ground spring attached

It is possible to make your own probe ground spring using TCW25 tinned copper
wire, see (Figure 2). Form the spring around the probe, remove and apply a little
flux and solder to hold it in place. Trim the ends and use appropriately.

3 Oscilloscope and probes
When measuring fast switching nodes always use a high-quality oscilloscope that
does not introduce unwanted parasitic capacitance, especially in the sensing loop.

Figure 5 | Typical oscilloscope used in a Nexperia measurement lab

To make accurate measurements, the best quality scope probes should be used.
These should have high input impedance: preferably 100 MΩ or greater, and very508

low capacitance: less than 4 pF. Also, the greatest bandwidth scope probes should
be used in line with the switching edges that are being measured.
The impedance and capacitance of the scope probe can have an effect on thevoltage that is being measured. By using high input impedance and very low
capacitance the effect of the probe on the node being measured is minimised.
Always check that the oscilloscope probe you are using has the correct maximum
voltage rating for the signal being measured. Do not exceed the manufacturer's
recommendations - voltages close to a probe’s maximum rating should be avoided.
Detailed information about probes and advice on measurement best practice isoften available from probe manufacturers. As an example, Tektronix publish acomprehensive guide - ABCs of Probes Primer, (http://info.tek.com/www-abcs-ofprobes-primer.html).

4 Probe positioner
When measuring high voltages extreme caution should be taken and best practices
adhered to.
With this in mind, the use of a probe positioner is highly recommended. The
oscilloscope probes can be adjusted to make contact with the required
measurement points, leaving the test setup to be operated “hands free”.

22
Probing considerations for fast switching applications

MOSFET & GaN FET Application Handbook

The following probe positioner is recommended: Type : MSA100 3D PROBEPOSITIONER Farnell part number 1552771.

Figure 6 | Probe positioner

509

Probing considerations for fast switching applications

22

510

Nexperia | A Power Design Engineer's Guide

FiguresFigure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.

Switching waveform; probe grounding method comparison................ 506
Probing a fast switching node..................................................................... 507
Probe ground spring..................................................................................... 507
Oscilloscope probe with ground spring attached.................................... 508
Typical oscilloscope used in a Nexperia measurement lab..................... 508
Probe positioner............................................................................................ 509

Abbreviations

Abbreviations

MOSFET & GaN FET Application Handbook511

Abbreviations

Nexperia | A Power Design Engineer's Guide

512

2DEG
AC
AEC-Q101
AN
BJT
BW
C
Cadd
CGD
CGS
Ci
CISPR
CISS
CLK
COSS
Cp
CSP
Cth
DC
de-Q
dIs/dtDUT
EDS(AL)R
EMC
EMI
EOS
Eoss
EOX
ESD
ESL
ESR
FET
FIT
FR4
FRING
GaN
HEMT
HS
I/O
I2C-busID

2-Dimensional Electron Gas
Alternating current
Automotive Electronics Council Quality System
Application Note
Bipolar Junction Transistor
Bandwidth
Capacitor
Additional external capacitance fitted drain to source
Gate-drain capacitance
Gate-source capacitanceConstituent thermal capacitance element
International Special Committee on Radio Interference
Input capacitanceParasitic capacitance
Output capacitanceSpecific heat capacity
Chip Scale Package
Thermal capacitanceDirect current
Reduce the quality factor of a resonance; damp
Time rate of change of source current
Device Under Test
Non-repetitive drain-source avalanche energy
Electro-magnetic compatibility
Electro-magnetic interferenceElectrical Overstress
Energy stored on output capacitance
Oxide electric field strength
ElectroStatic Discharge
Equivalent series inductance
Equivalent series resistanceField Effect Transistor
Failure In Time
Flame Retardant epoxy glass laminate (see NEMA LI 1-1998)
Oscillation (ringing) frequency
Gallium Nitride
High Electron Mobility Transistor
High side MOSFET
Input/Output
Inter-Integrated Circuit bus
MOSFET drain current

IDM
IDS(AL)S
IDSS
IEC
IGSS
IPC
IS
ISM
JEDEC
KGD
L
LFPAK
LLK
LS
LV
MOSFET
NTC
OEM
P(t)
PCB
PSMN
PTC

Pulsed drain current
Non-repetitive drainsource avalanche current
Drain current source shorted (drain leakage current)International Electrotechnical Commission
Gate current source shorted (gate leakage current)
Institute of Printed Circuits
Source current
Peak source current
Joint Electron Device Engineering Council
Known Good Die
Inductor
Loss Free Package (Nexperia package naming)
Stray (leakage) inductanceLow side MOSFET
Low Voltage
Metal-Oxide Semiconductor Field-Effect Transistor
Negative Temperature Coefficient
Original Equipment Manufacturer
Power as a function of time
Printed Circuit Board
Power Silicon Max N-Channel
Positive Temperature Coefficient
Maximum continuous power the device can dissipate with the
Ptotmounting base held continuously at 25 °C
Pulse Width Modulation
PWM
Gate Charge
QG
Gate-Drain Charge
QGD
Charge stored in the output capacitance
QOSS
ResistorR
On-state bulk-drain-drain resistance
RBDD(on)On-state bulk-drain-source resistance
RBDS(on)Resistor-Capacitor
RC
DC Link RC snubber
RCDCL
Switching Node RC snubber
RCSN
Drain-source on-state resistanceRDS(on)Gate resistance external to the package
RG(ext)Constituent thermal resistance element
Ri
Load resistance
RL
Reverse polarity protection
RPP
On-state source-source resistance
RSS(on)Thermal resistanceRth
Rth(j-amb) / Rth(j-a) Thermal resitance (junction to ambient)Abbreviations

MOSFET & GaN FET Application Handbook513

Abbreviations

Nexperia | A Power Design Engineer's Guide

514

Rth(j-mb)
SN
SOA
SPICE
T
t
Tamb
Tc
TDDB
ti
Tj
Tj(rise)
Tmb
TOX
tp
Ts
Tstg
UIS
VBR
VD
VDD
VDGR
Vdrop
VDS
VG
VGS
VGS(th)
Vin
VS
WLCSP
ZTC
Zth(j-mb)
Zth(t)
δfactorΔT
ΔTj
ΔTj(max)
ΔTon
ζ

Thermal resistance (junction to mounting base)Switch node
Safe Operating Area
Simulation Program with Integrated Circuit Emphasis
Total time of heating pulse
RC time constantAmbient temperature
Case temperature (plastic housing)Time Dependant Dielectric Breakdown
Thermal time constantJunction temperature
Junction temperature rise in the MOSFET
Mounting base temperature of the MOSFET
Oxide thickness
Pulse width, pulse duration
Surface temperature
Storage Temperature
Unclamped Inductive Switching
Breakdown voltage
Drain to ground voltage
Application supply voltage
Gate-source voltage
Voltage drop
Drain-source voltage
Gate to ground voltage
Gate-source voltage
Gate-source threshold voltage
Input voltage
Source to ground voltage
Wafer Level Chip Scale Package
Zero temperature coefficient
Thermal impedance (junction to mounting base)Transient thermal impedance
Duty factor, duty cycle
Change in temperature
Average temperature rise from average
Maximum junction temperature variation
On-state temperature difference
Zeta, damping factor

IndexMOSFET & GaN FET Application Handbook

Index

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

515

IndexNexperia | A Power Design Engineer's Guide

A
Active clamping.............................................................................................................. 98–99
AEC-Q101 (Automotive Electronics Council Quality System)
GaN FETs..............................................................................................470, 471, 474, 485
MOSFETs...................................................................... 26, 64–65, 64, 67, 338–339, 373
All-Electric Vehicles (xEVs)....................................................................................... 478–479
Avalanche current..................................................... 35, 97, 104, 355, 356, 358, 359, 381
repetitive........................................................................... 109, 109, 110, 111, 357, 360
single-shot................................................................ 105–106, 107–108, 107, 108, 357
Avalanche energy............................................................................................................... 358
non-repetitive drain-source...................................... 30, 32, 35–36, 35, 104, 106, 107
repetitive drain-source.....................................................32, 35–36, 35, 109, 111, 359
Avalanche power........................................................... 105–106, 107, 109–110, 110, 113
Avalanche ruggedness............................................................................................. 354–361
repetitive.................................................................. 108–109, 109, 110, 357, 359, 360
limiting values......................................................................................... 32, 35–36, 35
safe operation............................................................. 110–111, 111, 112–113, 361
single-shot........................................................104–106, 105, 107, 113, 357, 359–360
limiting values.................................................................................. 30, 32, 35–36, 35
safe operation...................................................................... 107–108, 108, 112, 361
Avalanching
GaN FETs............................................................................................................... 453–454
MOSFETs..................................................................................... 72–73, 74, 97, 283, 378
B
Bare die (KGD) MOSFETs......................................................................................................96
Bidirectional bus systems................................................................................444–446, 445
Bipolar Junction Transistors (BJTs)............................ 354–356, 354, 358, 363, 364, 365
Body diodes
GaN FETs..................................................................................... 453, 464, 465, 490–491
MOSFETsdata sheets.................................................................... 27, 28, 29, 30, 32, 43, 47, 48
FAQs about data sheets...................................................... 347–349, 348, 363–364
half-bridge configuration........146, 147, 151–152, 154–155,158, 166, 349, 368
see also freewheeling diodes; schottky diodes; zener diodes
Bootstrap supply........................................................................................................315, 494
Breakdown voltage (VBR).................................................. 35, 52, 105–106, 109, 212, 381
see also drain-source breakdown voltage (V(BR)DSS);
gate oxide breakdown voltage
Buck converter................................................................................ 304, 306, 306, 314, 322
Bypass capacitors.....................................................................................159–160, 160, 496

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

516

C
Capacitances
bypass...................................................................................................159–160, 160, 496
capacitance characteristics
GaN FETs....................................................................... 459–463, 460, 461, 465–466
MOSFETs....................................................................... 43, 44–47, 47, 384–385, 385
DC-to-DC conversion..........................................................................426–428, 427, 494
drain-source capacitance (CDS). ............................. 47, 71, 71, 72, 151, 153, 155, 161, 161, 377, 378, 384–385, 430
heat sink......................................................................................................................... 500
junction capacitance............................................................................................155, 494
oscilloscope probes............................................................................................. 508–509
parasitic capacitance...................................................... 71–72, 71, 137–142, 429, 508
reverse transfer........................... 43, 46, 47, 161, 165, 384–385, 385, 459–460, 460
thermal capacitance................................................ 120–121, 120, 217, 219, 219, 346
see also gate-drain capacitance (CGD); gate-source capacitance (CGS)
Cascode configuration.......................................................... 459, 471, 480–482, 482, 501
Cauer RC thermal models..................................................... 118, 121–123, 122, 123, 271
CCPAK see Copper Clip bond Packages
Cell pitch....................................................................................................176, 178–183, 375
CFD series superjunction Si MOSFETs....................................................................491, 492
Chip Scale Packages (CSP)......................................................................437–438, 437, 438
CISPR requirements (International Special Committee on Radio Interference)..... 166
Computational Fluid Dynamics (CFD) simulation........................................340–341, 393
Conduction (heat)
GaN FETs ......................................................................................................479, 490, 495
MOSFETs......212, 213, 214, 214–215, 214, 218, 222, 288–290, 290, 346, 373–374
Continuous drain current
applications.......................................................................................................... 278–283
calculating maximum....................................................... 272–276, 272–274, 273, 275
data sheets.......................................................................................... 32–33, 33, 36, 179
FAQs about data sheets.................................................. 349, 352, 387–388, 387, 388
Continuous source current, maximum...................... 266, 276–278, 277, 278, 280, 282
Convection....................................................213, 215–216, 215, 215, 222, 288–289, 290
Copper clip bond packages
GaN FETs........................................................................................................................ 474
MOSFETs............................................................................ 266–269, 267, 268, 291, 297
see also LFPAK MOSFETs
Crystal oscillator and frequency divider................................................................308, 308

Index

MOSFET & GaN FET Application Handbook

D
D-mode (depletion mode) GaN FETs........................................... 470–474, 480–481, 490
D2PAK................................................................................... 90, 96, 261, 266, 267, 344, 345
Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

517

IndexNexperia | A Power Design Engineer's Guide

Data sheet parameters. GaN FETs..................................................................................................................... 466
dynamic characteristics.............................................. 459–465, 460, 461, 463, 464
graphic symbols.....................................................................................451–452, 452
limiting values.........................................................................................452–456, 453
nomenclature..................................................................................................450, 450
pinning information......................................................................450–451, 450, 451
recovered charge............................................................................................465, 465
static characteristics.....................................................................457–458, 457, 458
switching.................................................................................................................. 466
MOSFETsderating curves.........................................................................32–36, 33–35, 55, 55
dynamic characteristics...............................................43–48, 43, 44, 46, 47, 48, 51
limiting values................................................................................................29–36, 30
ordering information................................................................................................29
pinning information............................................................................................ 29, 29
power dissipation..........................................................................................32–34, 34
product overview................................................................................................ 26–27
quick reference data....................................................................................27–28, 27
Safe Operating Area (SOA) curves......................................... 36, 36, 49–55, 49, 55
static characteristics........................................................ 38–42, 38, 39, 41, 42, 273
thermal characteristics..................................................... 36–37, 37, 37, 54–55, 55
see also derating curves; power dissipation; Safe Operating Area (SOA) curves
DC-link snubbers......................................................................................466, 466, 493, 496
DC-to-AC (traction) inverters............................................... 470, 480, 483, 483, 484, 485
DC-to-DC conversion...................................................70, 79, 92, 146, 279, 279, 281, 372
DC-to-DC down-converters .........414–420, 415, 416, 418, 419, 423–428, 425, 427
DC-to-DC up- and down-converters.........................................................422–423, 423
DC-to-DC up-converters....................................................................420–422, 421, 422
GaN FETs.......................................................................................................483, 483, 494
switching losses......................................414–415, 417, 419, 420, 428–431, 429, 430
see also H-bridge motor controller
Depletion mode (D-mode) GaN FETs........................................... 470–474, 480–481, 490
Derating curves
GaN FETs........................................................................................................................ 452
MOSFETs........................32–36, 33–35, 49, 49, 55, 55, 274–275, 275, 349–350, 350
dI/dt (time rate of change of current)......147, 157–158, 162, 163–164, 164, 170, 484
dID/dt (time rate of change of drain current)...............................................152, 154, 155
Diodes see Body diodes; Freewheeling diodes; Schottky diodes; Zener diodes
Discontinuation of Delivery (DOD) document.............................................................. 376
Double pulse testing
GaN FETs........................................................................................................................ 495
MOSFETs....................... 146–156, 147, 149, 150, 154, 165, 165, 168–169, 169, 170
DPA........................................................................................................................ 96, 154, 261
Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

518

Drain current (ID)GaN FETs ......................................................................................................453, 456, 495
power MOSFETs
data sheets.......................................................................... 27, 28, 30, 31–33, 39, 48
drain-source current path.....................................................................291, 291, 295
FAQs about data sheets...................................................... 383, 384, 390, 391, 391
paralleled groups................................................................................... 85, 89, 93, 96
small-signal MOSFETs........................................................................409, 430, 439–440
see also continuous drain current; dI/dt (time rate of change of current); dID/dt(time rate of change of drain current); double pulse testing; drain leakage
current (IDSS); peak drain current (IDM)
Drain-gate voltage (VDGR)............................................................................................. 31, 45
Drain leakage current (IDSS).............................................. 38, 40, 266, 380, 380, 381, 457
Drain-source breakdown voltage (V(BR)DSS)
avalanche events..................................................... 105–106, 107, 109, 109, 212, 404
data sheets........................................................................................ 38, 39, 52, 381, 383
drift engineering.......................................................................................................... 375
gate-drive operation................................................................................................ 73, 74
paralleled groups..................................................................................................... 97–99
Drain-source capacitance (CDS). ................................... 47, 71, 71, 72, 151, 153, 155, 161, 161, 377, 378, 384–385, 430
Drain-source leakage............................................................. 266, 402–404, 403, 407, 409
Drain-source on-state resistance (RDS(on))
GaN FETs...........................................................457–458, 458, 470, 472, 472, 479, 485
power MOSFETs
data sheets........................... 27, 28, 38, 40–42, 41, 42, 49, 50,151, 273, 273, 273
FAQs about data sheets...................................................... 338, 372–375, 387, 387
gate oxide thickness.........................................................................................66, 338
gate-source voltage...........................28, 41, 41, 60–61, 65, 69, 69, 151, 273, 338
paralleled groups.........................................................85–86, 87, 88–89, 89, 93–94
small-signal MOSFETs...................408–409, 409, 423, 428–431, 437, 438, 439–440
Drain-source voltage (VDS)GaN FETs................................................................... 450, 453–456, 453, 460, 460, 485
power MOSFETs
data sheets...................... 27, 28, 29–30, 30, 36, 40, 45–46, 46, 47, 51–52, 55, 55
FAQs about data sheets.................................... 352, 382, 385, 387, 387, 388, 391
paralleled groups...................................................................................85, 87, 88–89
voltage spikes.............................................152, 156–159, 159, 164, 175, 349, 363
small-signal MOSFETs................... 403, 403, 404, 408, 409, 430, 430, 439–440, 444
see also drain-source breakdown voltage (V(BR)DSS); ringing
Drift engineering..............................................................................................374–375, 375
Duty cycle generator.................................................... 303, 307–310, 307, 309, 309, 323
dV/dt (time rate of change of voltage)

Index

MOSFET & GaN FET Application Handbook

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

519

IndexNexperia | A Power Design Engineer's Guide

GaN FETs..............................................................................................471, 478, 484–485
MOSFETs..72, 72, 161, 161, 163–165, 163, 164, 165, 170, 362–366, 365, 368, 386
Dynamic characteristics
GaN FETs............................................................................ 459–465, 460, 461, 463, 464
MOSFETs .................................................................................. 43–48, 43, 44, 46, 47, 48
E
E-mode (enhancement mode) GaN FETs.............................................470–471, 480–481
EEHE conference................................................................................................................ 113
Electric Vehicles (EVs).......................................................................................478–479, 482
Electrical Overstress (EOS)...................................................................................... 174–205
Human Body Model (HBM) ESD............................ 175, 175, 176, 183–188, 184–188
linear mode operation. .....................................179–180, 179, 180–181, 195, 195–203, 197, 199, 201, 203
over-current............................................ 181–182, 182–183, 182, 204–205, 204–205
Unclamped Inductive Switching (UIS)......... 177–178, 177, 178, 189–194, 189–194
Electrical resistance........................................................................ 120, 120, 217–218, 218
Electrical–thermal analogy.......................................... 120, 120, 217–220, 217, 218–220
ElectroMagnetic Compatibility (EMC)...........................................................146, 156, 166
see also Half-bridge switching
ElectroStatic Discharge (ESD).............................376–380, 377, 379, 402, 402, 405, 408
Human Body Model (HBM).................................... 175, 175, 176, 183–188, 184–188
Enclosure design........................................................................................................289, 289
Enhancement mode (E-mode) GaN FETs.............................................470–471, 480–481
EOS see Electrical OverstressF
Failure In Time (FIT) rates.......................................................................394–395, 395, 485
Failure modes......................................................................... 357–359, 363, 471, 472, 473
Failure signatures...................................................................................................... 174–205
Human Body Model (HBM).................................... 175, 175, 176, 183–188, 184–188
linear mode operation. .....................................179–180, 179, 180–181, 195, 195–203, 197, 199, 201, 203
over-current............................................ 181–182, 182–183, 182, 204–205, 204–205
Unclamped Inductive Switching (UIS)......... 177–178, 177, 178, 189–194, 189–194
Fall time.............................................................................................. 43, 164, 166, 369, 370
Ferrite beads........................................................................... 463, 464, 494, 495, 495, 501'Flotherm' simulations.....................................................................................246, 258, 346
Foster RC thermal models.................................................... 118, 120–122, 120, 121, 122
Freewheeling current, GaN FETs....................................................................465, 471, 490
Freewheeling diodes......................................................... 97, 98, 415, 429, 490–491, 494
Frequency selection jumpers.................................................................302, 307, 308, 322
Full bridges see H-bridge motor controller; H-bridge motor drive

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

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G
Gallium Nitride (GaN) properties..................................................................................... 480
GaN FETs
half-bridge configuration.............472, 474, 484, 490–503, 490–493, 495, 496–501
Power GaN technology.................................. 470–474, 474, 478–486, 480, 481–484
GaN HEMT (High Electron Mobility Transistor).470, 480–482, 481, 482, 484, 490, 501
Gate bounce
E-mode GaN FETs.................................................................................................471, 480
MOSFETs.............................................................72, 161, 163, 366, 367–369, 368, 369
Gate charge (QG)GaN FETs..............................................................................................430, 459, 480, 494
MOSFETs..............................................43, 44–46, 44, 46, 48, 373, 374, 379, 390, 390
Gate-drain capacitance (CGD)
data sheets.....................................................................................................47, 384–385
dV/dt.............................. 161, 161, 164–165, 164, 165, 362, 364, 365, 367, 367, 368
gate-drive operation................................................................................... 71–72, 71, 72
gate oxide.....................................................................................................376, 377, 378
half-bridge configuration.............150–151, 151, 153, 155, 161, 161, 164–165, 165
Gate-drain charge (QGD) see Miller plateau (gate-drain charge (QGD))
Gate-drain resistance.......................................................................................164–165, 165
Gate drive
GaN FETs...........................................................471, 480, 492, 494, 495, 495, 501, 502
power MOSFETs
capacitances......................................150–151, 154–155, 157, 161, 161, 163–164
external....................................................................................................................... 68
external resistance..........................147–148, 161–163, 161–163, 368–369, 369, 369, 370, 370, 386
gate-source threshold voltage........................................ 40, 61–62, 161, 381, 382
gate to ground.................................................................................68–75, 69, 70, 73
gate to source..................................................................................68, 69, 71, 74, 74
paralleled groups................................................................................................ 95–96
small-signal MOSFETs........................................................................430, 435, 437, 463
Gate ferrite beads.................................................................. 463, 464, 494, 495, 495, 501
Gate (input) loop........................................................... 451, 451, 496, 496, 501–502, 501
Gate leakage current (IGSS)..................................................................38, 40, 266, 408, 457
Gate oxide.................................................................................... 31, 65–67, 65, 68, 78, 338
avalanching.................................................................................................................... 360
capacitances.................................................................................................376–379, 384
failure signatures..................................................................................................175, 176
Gate oxide breakdown voltage.............................................................376–377, 379, 405
Gate resistance........................................................................38, 42, 47, 95, 168, 168, 366
external/gate resistors. ........43, 95–96, 147–148, 161–163, 161–163, 317, 321, 368–370, 369, 369, 370,
370, 386

Index

MOSFET & GaN FET Application Handbook

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GaN FETs.......................................................................................................494, 495, 495
Gate signal feedback........................................................................................161–162, 161
Gate-source capacitance (CGS)
channel length.............................................................................................................. 375
data sheets..............................................................................................................47, 384
DC-to-DC conversion........................................................................................... 429–430
dV/dt.........................................................161, 163, 164, 362, 364, 365, 367, 367, 368
gate-drive operation..........................................................................................71–72, 71
half-bridge configuration....................................... 150, 151, 153–155, 161, 163, 164
Gate-source charge (QGS)..........................................28, 43, 44–46, 44, 46, 363, 459, 463
Gate-source ESD rating....................................................................................378–379, 379
Gate-source leakage....................................379, 402, 402, 404–405, 405, 407–409, 409
Gate-source threshold voltage (VGS(th))
GaN FETs............................................................................ 457, 457, 471, 480, 490, 491
power MOSFETs
data sheets.................................................... 38, 39–40, 39, 44–46, 51–52, 51, 352
dV/dt...............................................................................................161, 362, 367–369
gate drive operation..............................................................................60–62, 61–62
half-bridge configuration................................................... 151, 154, 161, 367–369
low temperature......................................................... 381, 382, 383, 385–386, 386
paralleled groups..........................................................................................94, 98–99
plateau voltage......................................................... 45–46, 61, 390–391, 390, 391
Spirito boundary............................................................................................ 351–352
small-signal MOSFETs...................................................... 408, 429, 430, 431, 445–446
Gate-source voltage (VGS)GaN FETs............................................................................ 453, 455–456, 490–491, 491
power MOSFETs
data sheets.............................28, 30, 31, 41, 41, 45–47, 46, 48, 51, 51, 60–63, 63
drain source on-state resistance.....28, 41, 41, 60–61, 65, 69, 69, 151, 273, 338
FAQs about data sheets.................................... 338–339, 351, 362, 378, 383, 384
small-signal MOSFETs............................................................... 409, 430, 430, 445–446
see also double pulse testing; gate bounce; mission profiles; plateau voltage
Gate to ground voltage.............................................................................68–75, 69, 70, 73
Ground trace/plane.................................................................................496–497, 500, 502
HH-bridge motor controller............................................................. 302–304, 302, 303, 333
bill of materials.................................................................................................... 326–333
clock........................................................................... 307–309, 308–310, 309, 322–323
drive circuits............................................................................... 313–315, 313–315, 325
H-bridge circuit.......................................................................... 316–318, 316–317, 324
over-current detection......................................................................318–319, 319, 324
power supply....................................................................... 76, 304–306, 304–307, 322
Pulse Width Modulation (PWM) circuit.......310–313, 310–313, 318, 320, 320, 323
Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

522

H-bridge motor drive........................................................................................75–78, 75–77
Half-bridge switching
double pulse testing. ............................ 146–156, 147, 149, 150, 154, 165, 165, 168–169, 169, 170, 170
GaN FETs.........................................472, 474, 484, 490–503, 490–493, 495, 496–501
gate bounce................................................................72, 161, 163, 366–367, 366, 367
switching performance improvements........................ 156–166, 161–162, 164–166
Heat propagation see Conduction (heat); Convection; Radiation
Heatsinking........................................................................................................ 52–53, 85, 92
copper clip bond package........................................................ 290, 291, 291, 345, 389
LFPAK33................................................................................ 229, 230, 235, 237, 237
LFPAK56................................................................................ 248, 249, 251, 253, 253
LFPAK56D.............................................................................. 225, 225, 231, 233, 233
LFPAK88.............................................248, 249, 251, 253, 253, 267, 267, 270, 346
failure signatures..................................................................................................178, 180
GaN FETs.......................................................................................................496, 500–501
infinite heatsink............................................................................................... 88–89, 392
High Electron Mobility Transistor (HEMT)........470, 480–482, 481, 482, 484, 490, 501
High-Temperature Gate Bias (HTGB) test..............................................................395, 473
High-Temperature Operating Life (HTOL) tests.................................395, 472–473, 473
High-Temperature Reverse Bias (HTRB) test......................................395, 472, 472, 473
High-Temperature Storage Life (HTSL) test.................................................................. 395
Hot spotting (Spirito effect)..........................................54, 351, 352–353, 352, 354, 355
Human Body Model (HBM).........................175, 175, 176, 183–188, 184–188, 377, 379
HV isolation tests...................................................................................................... 371–372

Index

MOSFET & GaN FET Application Handbook

I
IEC 60134 rating systems for electronic tubes and valves and analogous
semiconductor devices.................................. 29, 30, 63, 272, 274, 277, 452–456, 453
IEC 61000-4-2 ElectroStatic Discharge (ESD) immunity test...................................... 380
IGBTs (Si Insulated-Gate Bipolar Transistors)......................................470, 479, 483, 490
Impedance................................................................................72, 91, 93–94, 152, 163, 496
oscilloscope probes............................................................................................. 508–509
see also thermal impedance (Zth)Inductancesavalanching.................................................................................................................... 283
DC-to-DC conversion........................................................ 415, 416, 418–420, 426–427
external inductances......................................................................................................28
GaN FETs................................452, 484–485, 490, 492–494, 496–497, 497, 499–501
loop inductance.............................152–153, 153, 154–155, 155–158, 157, 159, 501
measuring current........................................................................................................ 148
paralleled groups...............................................................................................93–94, 93
parasitic (stray) inductance (LLK)GaN FETs..................................................... 484–485, 490, 492–493, 496–497, 497
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MOSFETs.............................................................. 137–142, 137, 138, 157, 177, 266
UIS tests......................................................................................................................... 177
Inductive current .........................................................................................................73, 368
Inductive energy dissipation........................................................................... 73, 97–99, 98
Inductive loads.................................................... 72–74, 73, 74, 97–98, 98, 367, 367, 389
see also load drive; load switches; Unclamped Inductive Switching (UIS)
Input capacitance (CISS)........... 43, 46, 47, 150–151, 151, 152, 154, 376–380, 384, 385
GaN FETs..............................................................................................459–460, 460, 501
Input (gate) loop...................................................451, 451, 463, 496, 496, 501–502, 501
Inter-Integrated Circuit bus (I2C-bus).............................................................444–446, 445
Inverters......................................................................... 146, 367, 407, 407, 470, 478–479
DC-to-AC/traction inverters............................................ 470, 480, 483, 483, 484, 485
IPC-2221 generic standard on printed board design................................................... 292
Isolators, using MOSFETs as.......................................................... 278, 279, 281–282, 281
J
JESD22-A114 ElectroStatic Discharge (ESD) sensitivity testing Human Body Model(HBM)................................................................................................................................ 377
JESD24-5 single pulse Unclamped Inductive Switching (UIS) avalanche test method............................................................................................................................................ 359
Jumpers.......................................................................... 302–303, 307, 308, 317, 321, 322
Junction capacitance.................................................................................................155, 494
Junction temperature (Tj)
GaN FETs..................................................................................... 453, 455, 456, 458, 458
power MOSFETs
calculations of ................................................................................................ 119–120
data sheets............................................................. 30, 31, 33, 39, 40, 41–42, 41–42
expected product lifetime......................................................68, 77–78, 77, 78, 97
FAQs about data sheets..........347–349, 356, 383–386, 384, 389, 392–393, 392
LFPAK33.................. 228–230, 230, 235, 237–238, 237, 239–241, 241, 245, 247
LFPAK56.............................................242–243, 243, 247–249, 249, 251, 253, 253
LFPAK56D
224–228, 225, 227, 228, 233–235, 233–234, 238–239, 239, 242–243, 243, 245
LFPAK88......................................................247–249, 249, 251, 253, 253, 255–256
small-signal MOSFETs.................................................................................404, 405, 408
see also RC thermal models; Safe Operating Area (SOA) curves
K
KGD (Known Good Die) MOSFETs .....................................................................................96
L
Lead-free (Pb-free) solder .............................................................................................. 373
Level shifting.........................................................310, 310, 312, 312, 323, 444–446, 445
LFPAK MOSFETs...........................................................................................................96, 261
copper clip bond package............................................... 266–269, 267, 268, 291, 297
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LFPAK33......................................................................................................................... 113
H-bridge motor controller.............................................................................303, 317
power dissipation
228–230, 229–230, 235, 236, 236–238, 237–238, 239–241, 240–241, 244–
245, 244–245, 246, 247
LFPAK56
H-bridge motor controller.............................................................................303, 317
power dissipation............243, 247–254, 248–250, 250, 252, 252–254, 254, 346
LFPAK56D...................................................................................................................... 261
H-bridge motor controller.............................................................................303, 317
power dissipation
224–228, 225–228, 231, 232, 232–235, 233–235, 238–239, 239, 242–245,
243–245, 343
LFPAK56E...................................................................................................................... 346
LFPAK88......................................................................................................................... 261
power dissipation..................................... 247–251, 248–250, 250, 252, 252–254,
253–258, 254, 256–257, 258, 259–260, 267, 267, 346
Life testing.............................................................64–65, 66, 67, 394–395, 471–473, 485
Linear mode operation
derating/safe operation......................................49, 49, 50, 52–55, 55, 353–354, 375
EOS tests/failure signatures. .....................................179–180, 179, 180–181, 195, 195–203, 197, 199, 201, 203
high-side MOSFETs with inductive loads.............................................................. 72–74
paralleled group design....................................................................................94–95, 95
thermal runaway ...................................................... 49, 49, 51–52, 51, 62, 69, 69, 351
Linear regulators............................................................................. 304, 306, 307, 322, 414
Load drive......................................................................................... 278–282, 279, 280, 281
see also inductive loads; resistive loads
Load switches.................................................47, 406–407, 406, 409, 434–440, 434–439
see also thermal resistance (Rth)Logic level MOSFETs. ................................... 41, 47, 61–62, 62, 63, 66, 78, 338–339, 362, 377, 444–446, 445
Loop inductance..................................152–153, 153, 154–155, 155–158, 157, 159, 501
Low leakage MOSFETs.....................................................................................406–409, 409

Index

MOSFET & GaN FET Application Handbook

M
Maximum power......................................................................................269–272, 272, 277
see also power dissipation
Miller plateau (gate-drain charge (QGD))
data sheets...................................................................................... 27, 28, 43, 44, 45, 46
GaN FETs................................................................................................................459, 492
Gate-drive design ......................................................................................61, 68, 95, 363
half-bridge configuration...........................................................................151, 163, 165
small-signal MOSFETs.................................................................................................. 430
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Mission profiles................................................................ 67–68, 68, 70, 72, 74–78, 75–78
MOSFETs, product overview........................................................................................ 26–27
Motor control........................................................................................................................79
see also H-bridge motor controller
Motor drive.......................................................................74, 261, 279–282, 279, 280, 281
H-bridge motor drive..................................................................................75–78, 75–77
Mounting base temperature
calculating maximum continuous current. .........................................................................267, 270, 274–276, 275, 387–389, 388
data sheets...............................................27, 30, 31–34, 33–34, 36, 40–41, 52–53, 55
FAQs about data sheets.............................................................................349, 350, 351
GaN FETs................................................................................................................455, 455
junction temperature........................................................................342, 392–393, 392
paralleled groups..................................................................................................... 90, 94
thermal simulation examples.................................................. 123–124, 129–130, 258
NN-channel MOSFETs................................29, 29, 409, 409, 415, 421, 421, 423, 445, 452
N-channel MOSFETs load switches......................................... 407, 407, 435, 435, 436
Negative Temperature Coefficient (NTC)...........................51, 62, 94, 99, 351, 352, 457
Non-repetitive drain-source avalanche current....... 105–106, 107–108, 107, 108, 357
Non-repetitive drain-source avalanche energy........... 30, 32, 35–36, 35, 104, 106, 107
O
Operating profiles........................................................... 67–68, 68, 70, 72, 74–78, 75–78
Ordering information................................................................................ 29, 326–333, 376
Oscillations......................................................................42, 47, 93, 95, 126, 153, 153, 155
methods of suppression.............................................................................................. 156
bypass capacitors..........................................................................159–160, 160, 496
GaN FETs..................................................................................................492–495, 501
gate driver resistance ...................................................................................164, 386
RC snubbers................................................136–138, 137, 138, 140, 142, 160, 161
Oscilloscopes.........................................................126, 148, 506–509, 506, 507, 508, 509
Output capacitance (COSS)GaN FETs................................................................... 459–462, 460, 463, 465, 497, 500
power MOSFETs
data sheets................................................................................... 43, 46, 47, 385, 385
half-bridge configuration.......................................... 153, 153, 155, 158–159, 159
RC snubbers...................................................................................137, 137, 138, 363
small-signal MOSFETs............................................................... 415, 418, 423, 426–427
Output charge (Qoss)................................................................28, 155, 461, 463, 463, 465
Output isolator MOSFETs................................................................................281–282, 281
Output load drive MOSFETs.......................................................... 278–282, 279, 280, 281
Output (power) loop...............................................................................451, 451, 496–500

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526

Over-current
EOS tests/failure signatures................ 181–182, 182–183, 182, 204–205, 204–205
H-bridge motor controller..................................... 302, 303, 304, 318–320, 319, 324
Half-bridge configuration .........................................................................366–367, 366

Index

MOSFET & GaN FET Application Handbook

P
P-Channel MOSFETs load switches. .................................. 406–407, 406, 409, 434, 434, 435–436, 437, 438–439, 438, 439
Packaging factors
GaN FETs..............................................................................................450–451, 450, 496
power MOSFETs .................... 96, 152, 154–155, 157, 371–374, 388–390, 391–392
data sheet......................................................................................................26, 29, 31
small-signal MOSFETs.................................................................................408–409, 409
see also Chip Scale Packages (CSP); copper clip bond packages; wire bond
MOSFETs
Paralleled group design................................................................................................ 84–99
dynamic operation.............................................................................................92–94, 93
gate drive considerations........................................................................................ 95–96
inductive energy dissipation.............................................................................97–99, 98
linear mode operation.......................................................................................94–95, 95
mounting and layouts...........................................................................90–92, 90, 91, 92
packaging considerations....................................................................................... 96–97
static operation........................................................................ 85–89, 87, 87, 88–89, 88
Parasitic BJT turn-on.................................................... 354–356, 354, 358, 363, 364, 365
Parasitic capacitance........................................................... 71–72, 71, 137–142, 429, 508
Parasitic (stray) inductance (LLK)GaN FETs.......................................................... 484–485, 490, 492–493, 496–497, 497
MOSFETs..................................................................................... 137–142, 157, 177, 266
Pb-free (lead-free) solder................................................................................................. 373
PCBs see Printed Circuit Boards (PCBs)
Peak drain current (IDM)GaN FETs................................................................................................................453, 456
MOSFETs......................................... 30, 31, 36, 55, 179, 182, 350, 352, 387, 387, 388
Peak drain-source avalanche power (PDS(AL)M).............................................105–106, 107
Peak soldering temperature, GaN FETs.................................................................453, 456
Peak source current (ISM)............................................................................ 30, 32, 453, 456
Pinning information
GaN FETs............................................................................ 450–451, 450, 451, 496, 496
MOSFETs........................................................................................................... 29, 29, 276
Planar technologies.................................................................................351, 354–355, 354
Plateau voltage
power MOSFETs....................................... 45–46, 46, 48, 61, 163, 390–391, 390, 391
small-signal MOSFETs......................................................................................... 429–431

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Positive Temperature Coefficient (PTC).................................................30, 51, 85, 89, 97
Power allowance (maximum power).............................................................269–272, 272
see also power dissipation
Power conversion, GaN FETs....................................... 470, 474, 478–479, 482, 485–486
Power dissipation
data sheets.................................................................. 27, 28, 30, 31, 32–34, 34, 50–52
design environment...........................................................................212, 220, 223, 260
EOS tests/failure signatures....................................................................................... 174
GaN FETs.......................................................................................................453, 455–456
junction temperature............................................. 342–343, 345, 347–349, 360, 383
LFPAK (all)...................................................................................................................... 270
LFPAK33.................................................................................................................246, 247
1-layer copper............................................ 228–230, 229–230, 239–241, 240–241
4-layers copper + vias................................................. 235, 236, 236–238, 237–238
comparison of 2 to LFPAK56D....................................................244–245, 244–245
LFPAK56
1-layer copper ......................................................................247–250, 248–250, 250
4-layers copper + vias................................................. 251–254, 252, 252–254, 254
comparison of 2 to LFPAK56D.............................................................242–243, 243
testing....................................................................................................................... 346
LFPAK56D
1 active MOSFETs in half bridge............................... 227–228, 227–228, 234–235
1-layer copper..................................................... 224–226, 225–226, 238–239, 239
4-layers copper + vias................................................. 231, 232, 232–235, 233–235
comparison to 2 x LFPAK33.................................................................242–243, 243
comparison to 2 x LFPAK56.................................................................242–243, 243
LFPAK88
1-layer copper.............................................................. 247–250, 248–250, 250, 259
1-layer copper (2 parts)....................................................... 255–256, 256–257, 260
4-layers copper + vias.......................251–254, 252, 252–254, 254, 257–258, 258
mission profiles...................................................................................................69, 69, 73
packages..................................................................................... 388–390, 391–393, 393
PCB layouts........................................................................ 288, 292–297, 292–296, 345
small-signal MOSFETs......................................................................................... 439–440
Power GaN technology....................................... 470–474, 474, 478–486, 480, 481–484
Power MOSFETs, product overview........................................................................... 26–27
Power (output) loop................................................................................451, 451, 496–500
Power plane.......................................................................................................406, 407, 497
Powertrain applications .............................................................................................52, 113
Printed Circuit Boards (PCBs)
DC-to-DC down-converters........................................................................423–424, 425
layout optimization
GaN FETs .................................................... 484–485, 490–503, 491–493, 496–501

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528

MOSFETs....................................................................... 260, 288–297, 292–296, 294
paralleled groups..........................................................................................91–92, 91
thermal paths...................................................221–223, 222, 260, 271, 389–390, 437
thermal resistance............................................... 85–86, 221–223, 260, 341, 344–345
see also H-bridge motor controller; loop inductance; packaging factors; power
dissipation
ProbesHall effect and Rogowski based................................................................................ 148
oscilloscopes..................................................................... 506–509, 506, 507, 508, 509
Pull-up resistors.................................................................................................444, 445, 445
Pulse Width Modulation (PWM) circuits
H-bridge motor controller.............................310–313, 310–313, 318, 320, 320, 323
Half-bridge configuration...................................................................................146, 166
paralleled group.......................................................................................... 92–94, 93, 95
simulations........................................................................................................... 373–374
small-signal MOSFETs.................................................................................................. 406
three-phase motor drive............................................................................................. 386
PWL (PieceWise Linear) files...........................................................................126, 127, 127

Index

MOSFET & GaN FET Application Handbook

Q
Q101 accreditation see AEC-Q101 (Automotive Electronics Council Quality System)
Quick reference data...............................................................................................27–28, 27
R
Radiation.......................................................213, 216–217, 216, 216, 222, 288–289, 290
RC snubbers
designing snubbers.......................................................... 136–143, 136, 137–139, 141
GaN FETs..............................................................................................466, 466, 493, 495
H-bridge.................................................................................................................316, 317
Half-bridge configuration..........................................................................159, 160, 161
RC thermal models.................... 118, 120–131, 120–125, 127–130, 131, 271, 348, 359
Recovered charge (Qr)
GaN FETs................................................................................................................465, 465
MOSFETs............................................................................................27, 28, 43, 348–349
Repetitive avalanche ruggedness....................................... 108–109, 109, 110, 357, 359
limiting values.............................................................................................. 32, 35–36, 35
safe operation.......................................................... 110–111, 111, 112–113, 360, 361
Repetitive drain-source avalanche current....................... 109, 109, 110, 111, 357, 360
Repetitive drain-source avalanche energy.........................32, 35–36, 35, 109, 111, 359
Repetitive drain-source avalanche power...........................................109–110, 110, 113
Resistance see Drain-source on-state resistance (RDS(on)); Electrical resistance; Gate
resistance; Thermal resistance (Rth)Resistive loads................................................................................................................ 44, 47
Resonant circuits see Ringing
Reverse current protected load switches............................................436–439, 437–439
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Reverse Polarity Protection (RPP)
power MOSFETs...................................................... 278, 282–283, 282–283, 305, 305
small-signal MOSFETs.................................................................................438–439, 439
Reverse recovery
GaN FETs...........................................................465, 479, 481, 481, 484, 490–491, 492
MOSFETs................................................43, 47, 79, 155, 166, 348–349, 363–364, 429
waveforms/oscillations........................................48, 136, 137, 142, 158, 168–169
Reverse transfer capacitance (Crss). ........................................................... 43, 46, 47, 161, 165, 384–385, 385, 459–460, 460
Ringing
GaN FETs................................................................... 493–494, 495, 496, 501–502, 501
MOSFETs............................................................................ 157, 159–161, 316–317, 317
Rise time............................................................................................. 43, 148, 164, 369, 370
S
Safe Operating Area (SOA) curves
data sheets...................................................................................... 36, 36, 49–55, 49, 55
EOS tests................................................................................................................179, 179
FAQs about data sheet.................350, 350, 352–354, 352, 359–360, 361, 387, 387
paralleled groups............................................................................................................94
repetitive avalanche ruggedness............................................ 110–111, 111, 360, 361
single-shot avalanche ruggedness....................... 107–108, 108, 112, 359–360, 361
Schottky diodes...................................306, 415–416, 416, 417, 419, 421–424, 421, 429
Si Insulated-Gate Bipolar Transistors (IGBTs)......................................470, 479, 483, 490
Si SuperJunction (SJ) technology............................................................................470, 479
Silicon capability.......................................................................................274, 275, 275, 276
Silicon Carbide (SiC) properties....................................................................................... 480
Silicon (Si) properties......................................................................................................... 480
Simulation Program with Integrated Circuit Emphasis (SPICE)
data sheets........................................................................................................... 373–374
DC-to-DC converter................................415–419, 416, 418, 419, 421–422, 421, 422
Half-bridge configuration... 149, 149, 160, 161, 165, 168–169, 168, 169, 170, 170
thermal models....118, 120, 123–126, 125, 127, 129, 131, 217, 218, 342, 348, 359
Single-shot avalanche ruggedness...................................... 104–106, 105, 107, 113, 357
limiting values....................................................................................... 30, 32, 35–36, 35
safe operation.......................................................... 107–108, 108, 112, 359–360, 361
Small-signal MOSFETs....................................................................................................... 261
DC-to-DC conversion
calculation of switching losses...................................................428–431, 429, 430
DC-to-DC down-converters.............414–420, 415, 416, 418, 419, 423–428, 425
DC-to-DC up- and down-converters....................................................422–423, 423
DC-to-DC up-converters...............................................................420–422, 421, 422
leakage behaviour............................................................ 402–411, 402, 403–407, 409
see also Inter-Integrated Circuit bus (I2C-bus); load switches
Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

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SOA curves see Safe Operating Area (SOA) curves
Soft start mode............................................................................................................94, 282
Source current (IS)data sheets................................................................................................................ 30, 32
GaN FETs................................................................................................................453, 456
maximum continuous............................................. 266, 276–278, 277, 278, 280, 282
Source-drain voltage (VSD)GaN FETs.......................................................................................................464, 464, 465
MOSFETs......................................................................43, 177, 179, 277–278, 277, 278
SPICE models see Simulation Program with Integrated Circuit Emphasis (SPICE)
Spirito effect (hot spotting)...........................................54, 351, 352–353, 352, 354, 355
Standard level MOSFETs.........................41, 61–62, 61, 63, 66, 377, 381–382, 385, 409
Static characteristics
GaN FETs..............................................................................................457–458, 457, 458
MOSFETs................................................................................. 38–42, 38, 39, 41, 42, 273
Steady state thermals..........................................217, 218, 219, 220–224, 220, 276, 278
data sheets..........................................................................................................37, 40, 53
thermal impedance................................................................... 118–119, 118, 269, 271
Storage temperature
GaN FETs................................................................................................................453, 456
MOSFETs........................................................................................................... 30, 31, 267
Stray (leakage) inductance (LLK) see Parasitic (stray) inductance (LLK)
Supply OR-ing....................................................................................................438–439, 439
Switching losses
DC-to-DC conversion..............................414–415, 417, 419, 420, 428–431, 429, 430
GaN FETs.......................................................... 479–480, 483–484, 493–494, 495, 500
power MOSFETs.......... 153–156, 158, 161–165, 161–165, 212, 373–374, 385, 390
see also reverse recovery
Switching nodes................................................................................................................. 157
capacitance.................................................................................................................... 500
current............................................................................................................................ 368
measurements....................................................................................506–508, 507, 508
switch node voltage..........................................72, 147–148, 151, 161, 164, 166, 465
switching-node snubber.............................................................................466, 493–495
Switching performance.............................................................43–47, 370, 470, 479–480
see also Half-bridge switching; Load switches
Synchronous rectifiers.....................................................................................165, 165, 368
SyncFET.......................................................................................................................... 146
System-on-Chip (SOC) solutions...................................................................................... 426

Index

MOSFET & GaN FET Application Handbook

T
Thermal capacitance (Cth)............................................ 120–121, 120, 217, 219, 219, 346
Thermal characteristics
data sheets............................................................................... 36–37, 37, 37, 54–55, 55
Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

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IndexNexperia | A Power Design Engineer's Guide

see also thermal impedance (Zth); thermal resistance (Rth)Thermal conductivity............................................................... 31, 214, 214–215, 289, 480
see also conduction (heat)
Thermal design.......................................................... 75–78, 212–213, 261, 267, 291, 291
see also conduction (heat); convection; power dissipation; radiation
Thermal–electrical analog........................................... 120, 120, 217–220, 217, 218–220
Thermal impedance (Zth)cell density..................................................................................................................... 359
comparison of MOSFETs.....................................................................................346, 347
data sheets......................................................... 36–37, 37, 50, 54, 340–341, 341, 353
maximum power..........................................................................................269–272, 271
RC thermal models.................................118–121, 118, 119, 123–125, 125, 128, 128
Thermal mass (Cth)........................................................ 120–121, 120, 217, 219, 219, 346
Thermal nomenclature..................................................................................................... 224
Thermal propagation phenomena see Conduction (heat); Convection; Radiation
Thermal resistance (Rth).................................................................................................... 270
data sheets...........................................................................36–37, 37, 37, 50, 340–345
drift engineering..................................................................................................375, 375
dual MOSFETs.......................................................................................................228, 228
junction to ambient (Rth(j-amb)). ............................ 221–224, 221, 222, 246, 288–297, 290, 291, 341, 343–346, 344
avalanche events............................................................................................ 109–110
paralleled groups......................................................................85, 86, 87–89, 88–89
junction to case............................................................................................................. 345
junction to mounting base (Rth(j-mb))
avalanche events..................................................................................................... 109
calculating maximum power allowance.............................................269–272, 272
data sheets....................................................................37, 40, 50, 53, 290, 342–345
dual MOSFETs.......................................................................................................... 393
paralleled groups..........................................................................................86, 86, 89
RC thermal models.........................................................................................118, 118
thermal design....... 220–221, 221, 245–247, 246, 257–258, 258, 259, 260, 346
mounting base to ambient............ 85, 245–246, 257–258, 260, 344, 371, 392, 393
paralleled groups................................................................................. 85–91, 86, 88–89
RC thermal models.................................................................... 118, 118, 120–121, 130
Thermal–electrical analogy...............................................................217–218, 217, 218
Thermal runaway in linear mode................................. 49, 49, 51–52, 51, 62, 69, 69, 351
Thermal transients.....................................................................................................219, 220
Thermal vias................ 91, 231–238, 232, 232–238, 236, 251–255, 252, 252–254, 371
Thermals, steady state see Steady state thermals
Three-phase bridges............................................................................................. 70, 71, 386
Time Dependant Dielectric Breakdown (TDDB)....................................66–68, 75, 77, 78
Traction inverters................................................................... 470, 480, 483, 483, 484, 485

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

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Transconductance.............................................................................................152, 351, 390
Transfer characteristics.................... 48, 48, 51, 152, 154, 374, 383, 384, 390, 390, 391
Transient drain-source voltage, GaN FETs...........................................453, 454, 454, 456
Transient overvoltage protection...........................................................................304, 305
Transient region.........................................................................................................220, 271
Transient voltages............................................................................................................. 492
Trench technologies
drift region.................................................................................................................... 375
dV/dt...................................................................................................................... 363–365
gate-oxide...............................................................................65, 65, 338, 376–377, 379
generation 2.................................................................................................................. 380
generation 3......................................................................................................... 340–341
generation 6............................................................................... 338, 340, 366, 374, 383
generation 9................................................................................................... 79, 113, 339
UIS issues......................................................................................................354–355, 354
Turn-off waveforms.....................................149, 150–153, 150, 157, 169, 170, 369, 495
Turn-on/Turn-off delay.................................................................... 43, 369–370, 369, 370
Turn-on waveforms............................................................... 154–156, 154, 168, 169, 495

Index

MOSFET & GaN FET Application Handbook

U
Unclamped Inductive Switching (UIS)avalanche ruggedness........................................................................................ 354–361
repetitive avalanche events...................................... 104, 108–113, 109–111, 361
single-shot avalanche events..........105–108, 107, 108, 112, 357, 359–360, 361
EOS tests/failure signatures......................... 177–178, 177, 178, 189–194, 189–194
V
Vias............................... 91, 231–238, 232, 232–238, 236, 251–255, 252, 252–254, 371
Voltage spikes...................................... 152, 156–159, 159, 164, 175, 349, 363, 444, 485
W
Waveforms see Turn-off waveforms; Turn-on waveforms
Wide Band-Gap (WBG) materials....................................................................470, 479, 480
Wire bond MOSFETs.......................................................... 50, 96, 154, 266, 275, 344, 345
EOS tests/failure signatures.................................. 180, 180–183, 182, 189, 195, 204Z
Zener diodes.................................................................................... 306, 326, 360, 404–405
Zero Temperature Coefficient (ZTC) point........................................ 48, 51, 52, 351, 352

Page numbers in bold denote tables, italics denote figures, and those underlined denote QR code links.

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536

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Notes

Notes

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NotesNotes

MOSFET & GaN FET Application Handbook

NotesNexperia | A Power Design Engineer's Guide

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MOSFET & GaN FET
Application Handbook
A Power Design Engineer’s Guide
Copyright © Nexperia
2020
www.nexperia.com
ISBN 978-0-9934854-7-3
All rights reserved.
No part of this publication may be reproduced or
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