J. Magn. Soc. Jpn., 41, 132-139 (2017)<Paper>Low permeability composite magnetic core transformer
with high coupling coefficient and its application
to PFM controlled quasi-resonant mode flyback-type DC-DC converter
K. Sato*,**, T. Sato**, M. Sonehara**, and H. Takeuchi**

Nagano Prefecture General Industrial Technology Center, 1-3-1 Osachi-katamacho, Okaya, Nagano 394-0084, Japan**

Faculty of Engineering, Shinshu Univ., 4-17-1 Wakasato, Nagano, Nagano 380-0928, JapanWe describe a low permeability composite magnetic core transformer with a high coupling coefficient and its
application to a PFM controlled quasi-resonant mode flyback-type DC-DC converter. To increase the coupling
coefficient between primary and secondary windings embedded in a carbonyl-iron powder/epoxy composite magnetic
core with a relative permeability of 6, three types of winding layouts were investigated by using a 2D-FEA numericalsimulation and the characteristics were confirmed experimentally. The primary and secondary windings that werelaid out adjacent to each other were the best choice as it had the highest coupling coefficient at over 0.99. By
applying the composite magnetic core transformer with this coupling coefficient to a 48 V input/12 V output PFM
controlled quasi-resonant mode flyback-type DC-DC converter, the maximum efficiency was 92.5% at an output
power of around 28 W, and an efficiency of 90.0% was obtained at an output power of 60 W.
Key words: low permeability magnetic core, carbonyl-iron powder/epoxy composite magnetic core transformer,
winding layout, coupling coefficient, DC-DC converter, efficiency
1. Introduction
Recently, SiC/GaN power devices have been applied
to switching power converters. It is because they makepossible to have good advantages such as lower ON
resistance, higher speed switching operation and higher
temperature operation than the Si-based power devices1), 2). By using the novel power devices, it is possible not
only to increase the power conversion efficiency of theconverter but also to downsize the magnetic devices
used in the switching power converters since theswitching frequency can be increased to MHz band ortens of MHz. However, it is considered difficult to use
the magnetic devices consisting of dust core or Mn-Zn
ferrite core for such the very high frequency switching
power converter. M. Harrison3) stated that the possible
magnetic core for MHz band or beyond 10 MHz power
conversion would be Ni-Zn ferrite core only.
The authors are developing a novel magnetic core
as a candidate substitute for the Ni-Zn ferrite core, forexample, the carbonyl-iron powder/epoxy composite
magnetic core was developed4) and was applied toswitching power converters5),6). In order to increase thevolume resistivity of the composite magnetic core body,the surface-oxidation for the carbonyl-iron powder wasvery
effective4).
Thoughthesurface-oxidized
carbonyl-iron powder/epoxy composite magnetic core
had a low relative permeability of about 6, the magnetic
loss was very small up to 10 MHz in frequency. By
applying the composite magnetic core inductor to a 1
MHz switching buck-type DC-DC converter using GaN
power device, a maximum power conversion efficiency of
93% was obtained5). It was also reported that acomposite magnetic core transformer with copper-cladpolyimide tape windings embedded in the composite

132

magnetic core could realize a coupling coefficient k >0.99 even when using such the low permeability
composite magnetic core6). Furthermore, by applying
the composite magnetic core transformer to a PWM(Pulse Width Modulation) controlled flyback-type
DC-DC converter, the maximum power conversion
efficiency was 91% at 400 kHz switching and 85% at 1
MHz switching6).
In general, the low permeability magnetic material
is not suitable for transformer core because of large
leakage inductance owing to the low coupling coefficient
between the windings. This study was done to develop
the composite magnetic core transformer with highcoupling coefficient and to apply it to a PFM (Pulse
Frequency Modulation) controlled quasi-resonant mode
flyback-type DC-DC converter7).
This paper describes an investigation on the designof low permeability composite magnetic coretransformer with high coupling coefficient on the basesof the two-dimensional finite element analysis (2D-FEA)
numerical simulation and experiments. In addition, aPFM controlled quasi-resonant mode flyback-type
DC-DC converter using the fabricated composite
magnetic core transformer is also shown.
2. Surface-oxidized carbonyl-iron powder/epoxy
composite magnetic core transformer
2.1 Properties of surface-oxidized carbonyl-iron
powder/epoxy composite magnetic coreAs the soft magnetic metal powder for thecomposite magnetic core, non-reduction carbonyl-iron
powder with a mean diameter of 1.6 m was used. Inorder to prevent the electrical contact between theadjacent powders in the composite magnetic core and to

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017

suppress the over-lapped eddy current loss, highresistive oxidized layer of several tens of nm wasintroduced on the surface of the powders by thermaloxidation in air at 200 degrees-C for 6h4). The composite
magnetic core was fabricated by the casting andpost-curing using the composite slurry with thesurface-oxidized carbonyl-iron powder and epoxy
precursor solution. Since the nonmagnetic epoxy resin
between the magnetic particles behaves as a magnetic
gap, the composite magnetic core is hard to bemagnetically saturated due to its low permeability andhas superior superimposed DC current characteristics
when using the magnetic devices, so that thetransformer core can be gapless.
Fig. 1 shows the static magnetization curve of thesurface-oxidized carbonyl-iron powder/epoxy composite
magnetic core measured by using a VSM (Riken Denshi
Co., Ltd.; BHV-55). The saturation magnetization was
about 1 T. Though the composite magnetic core hadnonmagnetic epoxy resin, it exhibited a high saturation
magnetization around 1 T, so it was considered that theenergy density could be increased and the transformer
core could be downsized.
Fig. 2 shows the frequency dependence of therelative complex permeability of the surface-oxidized
carbonyl-iron powder/epoxy composite magnetic coremeasured by using an impedance/material analyzer

Magnetization M [T]

1.5
1
0.5
0
-0.5
-1
-1.5
-800
-400
0
400
800
Applied magnetic field H [kA/m](Agilent; 4291B) and a coaxial waveguide. The real partof the complex relative permeability r’ was low as about
6, but the imaginary part r” was as small as 10-2 orderat 10 MHz or less. Although not measured below 1 MHz
in frequency, the relative permeability r’ of about 6 wasconsidered to be constant at hundreds kHz band and themagnetic loss r” was considered to be small enough at
same hundreds kHz band.
The volume resistivity of the surface-oxidized
carbonyl-iron powder/epoxy composite magnetic core
was about 100 m4), which was 100 times higher thanthat of Mn-Zn ferrite core. Therefore, it was consideredthat the eddy current loss of the transformer core could
be significantly suppressed at least up to 10 MHz.
2.2 Transformer structure
The output voltage of the quasi-resonant mode
flyback-type DC-DC converter is controlled by changing
both the on pulse width and the switching frequency
according to the output current7). As the output power
becomes larger, the on pulse width expands and theswitching frequency decreases. Also, the soft switching
operation can be realized by using the resonance of thedrain-source voltage of the MOSFET due to the
inductance of the transformer primary winding and thecapacitor connected in parallel to the MOSFET.
Therefore, in order to increase the switching frequency,
it is necessary to reduce the inductance of the primarywinding. Since the composite magnetic core underinvestigation has a high saturation magnetization andthe operating range ΔB of the transformer can beenlarged, even when the switching frequency decreases
under heavy load, the magnetic saturation occurs
hardly, which is due to the low inductance (lowpermeability composite magnetic core).
Fig . 3 sh o w s th e c o mposite magnetic coretransformer structure under investigation in this paper.
The transformer winding was embedded in thecomposite magnetic core with a 30 mm-square size andFig. 1 Static magnetization curve of surface-oxidized
carbonyl-iron powder/epoxy composite magnetic core.

Permeability r', r"

10
5

r'

1
0.5

r"
0.1
0.05
0.01
1M

10M
100M
Frequency f [Hz]1G

Fig. 2 Complex relative permeability of surface- oxidized
carbonyl-iron powder/epoxy composite magnetic core.

Fig. 3 Composite magnetic core transformer withembedded windings.

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017133

type-A
Primary winding
Secondarywinding

type-BPrimary
winding-1

Primary
winding-2

Secondarywinding

type-CPrimary winding
Secondarywinding-1

0

Secondarywinding-2

Cross-sectional
structure of winding

2

2

2

1

1

1

Fig. 4 Three types of winding layouts for composite
magnetic core transformers with embedded windings.
10 mm in thickness. A copper-clad polyimide tapehaving a 70 m thick, 5 mm wide copper conductor line
formed on a 100 m thick, 5.2 mm wide polyimide tape
was used as the primary and secondary winding.
In this paper, inductance of the primary winding
was designed so that the switching frequency was about
500 kHz when the converter output power of 60 W8). Fora reason of the fabrication process of the tape wound
coil, the number of turns was determined to be an evennumber, the primary winding turn was 8, and thesecondary winding turn was 4. Since the composite
magnetic core had a low relative permeability, winding
layout becomes more important than when using othermagnetic core materials having a high relative
permeability. In general, when the primary andsecondary winding were wound in an interleaved
structure, the coupling coefficient between the windings
could be increased9)-11). It was easy to wind in aninterleaved structure when the number of turns ofprimary winding and that of secondary windings wereequal. However, since the number of turns of primaryand secondary winding of the transformer discussed inthis paper was different, three types of transformers
having different winding layout as shown in Fig. 4 werestudied. In the cross-sectional view of windings in thelower figure of Fig. 4, “P-1” to “P-8” means from firstturn to eighth turn of the primary winding, respectively.
Similarly, “S-1” to “S-4” means from first turn to fourth
turn of the secondary winding, respectively.
Type-A transformer consisted of one primarywinding and one secondary winding. The secondary
winding was concentratedly laid out near the center ofthe primary winding when looking at the cross-sectional
structure of the windings. Therefore, the innermost
conductors of the primary winding and the secondary
winding were not adjacent to each other.
Type-B transformer consisted of two primarywindings and one secondary winding. The secondary
winding was sandwiched between two primary windings

134

and wound four turns. In order to make the primarywinding eight turns, the winding end of one of the two
primary windings and the winding start of the otherwere connected in series outside the magnetic core.
Hence, the innermost conductors of the primarywinding and the secondary winding were adjacent toeach other, and the secondary winding were distributedand evenly laid out between the primary winding whenlooking at the cross-sectional structure of the windings.
Type-C transformer consisted of one primarywinding and two secondary windings. After the firstsecondary winding was wound four turns with theprimary winding, the second secondary winding waswound continuously four turns with the primarywinding. In order to make the secondary winding four
turns, two secondary windings were connected inparallel outside the magnetic core. As shown in Fig. 4,
the innermost conductors of the primary winding andthe secondary winding were adjacent to each other andthe secondary winding of type-C were alternately andevenly laid out between the primary winding whenlooking at the cross-sectional structure of the windings.
The inductance and the resistance of the two secondary
windings were different because the lengths and innerdiameters were different. Hence, it was considered thatan imbalance in the current flowing of the twosecondary windings may occurs. The following 2D-FEA
numerical simulation and experiments took thisbehavior into account.
2.3 Fabrication procedure of composite magnetic coretransformer
The fabrication procedure of the composite
magnetic core transformer is as follows. An air-corewinding having an inner diameter of 17.5 mm was
fabricated. Air-core windings were covered withpolyimide tape to prevent contact between the winding
and the composite magnetic core material. The
composite
slurrywas
preparedbymixingsurface-oxidized carbonyl-iron powder and two
component epoxy precursor solution. The slurry wasfilled in a mold in which the winding was arranged andthen cured at 120 degrees-C for 5 h. Thereafter, the sizeof the composite magnetic core was adjusted bypolishing. As the lead wire from the winding inside thecomposite magnetic core, the same copper-cladpolyimide tape was used.
3. Operation analysis for composite magnetic coretransformer
3.1 Analysis model and conditions
In order to clarify the operation characteristics ofthe three types of transformers, 2D-FEA numericalsimulation (Ansys; Maxwell) was introduced. As shownin Fig. 3, the analytical region was set to be the righthalf of the transformer cross section because of thesymmetry of the device structure. The analytical modelfor deriving various parameters was a model rotated

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017

Table 1 Parameters for 2D-FEA numerical simulation
for three types of composite magnetic core transformers.

one time around c-c’ by post processing of the simulator.
For this reason, the transformer core of the analyticalmodel had a cylindrical shape with a diameter of 30 mm
and a thickness of 10 mm, and the outer shape wasdifferent from the fabricated transformer. However, it
was considered that there was little influence on theanalysis result since the magnetic circuit of thetransformer core had almost the same structure. The
2D-FEA
could
notaccurately
simulatethethree-dimensional structure of the transformer
including the influence of the lead wire etc. Therefore,
quantitative consideration was difficult. However, it
was considered sufficient to consider the relationshipbetween the winding layout of the transformer and its
operating characteristics under the same magnetic
circuit structural condition.
Table 1 shows the properties used for analysis. The
copper loss of the winding was calculated with taking
the losses due to the skin effect and the proximity effectinto account. The magnetic properties of the composite
magnetic core were regarded as linear, and themagnetic hysteresis was not considered. The core loss
was calculated from the imaginary part of the complex
permeability shown in Fig. 2 and set as the loss tangent
tan (=r”/r’) in the 2D-FEA numerical simulation.
Since the measurement variation of the imaginary partof complex permeability in the low frequency below 10
MHz was large, the value of tanat 10 MHz was also
used for the analysis below 10 MHz. Analysis was
carried out under the constant voltage source excitationby using “Maxwell Circuit Editor” function capable ofcoupled analysis of FEA and the circuit simulation, thecircuit diagram as shown in Fig. 5 was introduced, and

5
2.5
0

Meas. Calc.
: type-A: type-B: type-C1

0.1

Meas. Calc.
: type-A: type-B: type-C

0.01
1

Coupling coefficient k

Fig. 5 Relationship between 2D-FEA numericalsimulation for composite magnetic core transformer andequivalent circuit.

7.5

-2.5
10
Resistance Rs []Conductor line
(Copper)Insulator layer(Polyimide)

Constant
Permeability: r’ = 6 (const.)
Loss tangent: tan= 0.002 (const.)
Conductivity: 0.01 S/m (100 m)
Conductivity: 5.8 × 107 S/mThickness: 70 m, Width: 5 mm
Conductivity: 0 S/m
Thickness: 100 m, Width: 5.2 mm

Inductance Ls [H]

Material
Composite
magnetic core

10

0.995
0.99
0.985
Meas. Calc.
: type-A: type-B: type-C

0.98
0.975
100k

1M
Frequency f [Hz]10M

Fig. 6 Frequency characteristics of three types oftransformers.
the operation characteristics of the transformers were
simulated. In Fig. 5, a sinusoidal voltage source wasconnected to the primary side of the transformer, and a
resistive load was connected to the secondary side. The
input voltage V1 was set to 48 Vrms.
Since the measured value of the relative
permeability in Fig. 2 was a small signal characteristic
and tan was regarded as constant value, the calculated
power loss may be estimated to be smaller than theactual power loss. However, for the purpose ofcomparing the differences in characteristics due to thewinding layout, the essential problem was considered tobe small.
3.2 Electrical characteristics
Fig. 6 shows the equivalent series inductance,
equivalent series resistance and coupling coefficient of
the primary side of the transformer at the time of
no-load. In Fig. 6, the plots are measured values byusing an impedance/gain-phase analyzer (Agilent;
4194A) and the solid lines are calculated ones. The

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017135

Table 2 Measured and calculated DC resistance of eachwinding of transformers.

type-B
type-C

Measured26.6 m14.5 m28.8 m14.8 m27.4 m
7.2 m

Calculated
24.0 m12.1 m24.1 m12.0 m24.7 m
6.3 m

99
Efficiency [%]type-A

Primary
SecondaryPrimary
SecondaryPrimary
Secondary

100

98
97

Output voltage V2 [V]f =1 MHz: type-A: type-B: type-C

24.1

23.9

0

2
4
6
8
Output current I2 [A]10

Fig. 7 Analysis results of load current dependence of
output voltage for three types of transformers.
coupling coefficient was obtained by the secondary side
open/short circuit method.
As shown in Fig.6, the equivalent series inductanceof the fabricated transformer was about 4.6 Hregardless of the winding layout. The self-resonancefrequency of type-B and type-C was lower than that of
type-A because of the parasitic capacitance increase dueto the winding layout5).
Table 2 shows the measured and calculated DC
resistance of each winding of the transformers. As
shown in Table 2, the DC resistance of the primarywindings was almost equal regardless of the winding
layout. However, since two windings were connected inparallel, the DC resistance of the secondary winding oftype-C was almost half.
Although there was the difference between themeasured values and the calculated ones of eachparameter, this was considered to be due to thepositional deviation of the winding in the thickness
direction of the magnetic core5) and the influence of theparasitic parameters of the lead wire from the winding
inside the composite magnetic core not considered in theanalytical model. Also, since 2D-FEA was only magnetic
field analysis, the influence of self-resonance did notappear in the calculation results. However, thetendency of the electrical characteristics of the threetransformers generally agrees with the measured values
and the calculated ones, it was considered that it waspossible to compare the difference in characteristics dueto the winding layout.

136

0

2
4
6
8
Load current I2 [A]10

Fig. 8 Analysis results of load current dependence ofefficiency for three types of transformers.

24

23.8

: type-A: type-B: type-C

96
95

24.2

f = 1MHz

3.3 Load current dependences of transformer
Fig. 7 shows the calculated load current dependenceof the output voltage V2 of transformer. As shown in Fig.
7, the output voltage V2 of type-A exceeded 24 V in the
no-load and light load, which was higher than the value
calculated from the turn ratio of 8:4. It was considered
to be due to the low relative permeability of thecomposite magnetic core. Since the relative permeability
was low, the main magnetic flux was considered not
only to have the component in the composite magnetic
core body but also to include the leakage component in
the region near to the primary winding. As a result, theeffective inner diameter of the secondary winding wasconsidered to be enlarged and the inductance also
increased. This was equivalent to the increase of thenumber of turns of the secondary winding. Hence, it wasconsidered that the output voltage became higher thanthe value calculated from the turn ratio of 8:4. This
operation suggests that the operating point of thetransformer may unintentionally change from theoperating point designed based on the specifications of
the converter. On the other hand, in type-B and type-C,
since the innermost conductors of the primary winding
and the secondary winding were adjacent to each otherand there was almost no difference in inner diameter,
the output voltage was almost equal to the calculated
value from the turn ratio of 8:4.
The output voltage of type-A transformer decreased
more than the other two types of transformers. It wasconsidered because type-A had lower coupling
coefficient between primary and secondary winding
than the other two types of transformers, and its
leakage inductance was larger than the other two
transformers.
3.4 Power efficiency and loss
Fig. 8 shows the calculated power efficiency of thetransformer at 1 MHz. The maximum efficiency of 99%
or more in any transformer was obtained. Since the loss
due to the excitation current was mainly at no-load andlight load, the type-A transformer with the lowest

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017

frequency decreases with increasing load current, there
is a possibility that high efficiency can be maintained in
the wide load condition when the composite magnetic
core transformer is applied to such converter.

Power loss [W]

10

1

0.1

I2 = 0 A: type-A: type-B: type-C

Power loss [W]

0.01
10

: type-A: type-B: type-C

1

I2 = 4 A: type-A: type-B: type-C

0.1
10Power loss [W]I2 = 2 A

I2 = 6 A: type-A: type-B: type-C

1

I2 = 8 A

0.1
100k: type-A: type-B: type-C

I 2 = 10 A
: type-A: type-B: type-C

1M
10M
1M
10M 100k
Frequency f [MHz]Frequency f [MHz]Fig. 9 Analysis results of frequency dependence ofpower loss for three types of transformers.

equivalent series resistance of primary winding at 1
MHz had the highest power efficiency. On the other
hand, at heavy load, type-C with lowest secondary
winding resistance and highest coupling coefficient
between windings had the highest power efficiency.
Fig. 9 shows the calculated values of the frequency
dependence of the power loss of the transformer. The
frequency at the minimum power loss varied with theload current value, and the composite magnetic coretransformers under investigation in this study exhibitedthe minimum power loss obtained at 500 kHz to 2 MHz
according to the load current intending for the PFM
controlled quasi-resonant type flyback converter in thisstudy. The power loss in the low frequency increased
because the winding loss increased and the input power
factor decreased, which was considered to be owing to
the low inductance and the increase of excitation
current with lowering frequency. On the other hand, it
was considered that the power loss increased in the highfrequency due to the skin effect and proximity effect of
the windings.
With increasing load current I2, the frequency at
the minimum power loss shifted to the low frequency
side in any transformer. In the quasi-resonant mode
flyback-type DC-DC converter, since the switching

3.5 Magnetic flux density distribution in transformer
Fig. 10 shows the calculated magnetic flux density
distribution of the three types of transformers when thefrequency of 1 MHz and load current of 10 A. In Fig. 10,
the in-phase flux component with the exciting current
and 90-degrees phase-lag component are shown, where
the stored energy in the transformer corresponds to thein-phase flux component and the energy loss
corresponds to the 90-degrees phase-lag component.
Since the magnetic loss due to tan (=r”/r’) of thecomposite magnetic core was considered to be smallenough at 1 MHz, the energy loss was considered to bemainly due to the windings.
From upper figure in Fig. 10, it was found that thein-phase magnetic flux density with the exciting current
occurred mainly inside the winding and its spatial
distribution was almost same in three types of winding
layouts. On the other hand, as shown in the lower figure
in Fig. 10, 90-degrees phase-lag magnetic flux density
component shows a difference depending on the winding
layout. In the type-A, 90-degrees phase-lag magnetic
flux density distribution was generated between the
conductors where the primary winding and thesecondary winding were adjacent to each other. In
particular, the magnetic flux was also distributed in thecomposite magnetic core at the beginning and the end of
the secondary winding. Also in type-B, 90-degrees
phase-lag magnetic flux was generated between the
conductors where the primary winding and thesecondary winding were adjacent to each other, but no90-degrees phase-lag magnetic flux was generated in
the composite magnetic core. Type-C had little
90-degrees phase-lag magnetic flux in the composite
magnetic core and the 90-degrees phase-lag magnetic
flux density between windings was smaller than type-B.
It was considered that the 90-degrees phase-lag
magnetic flux density component was due to the energy
loss owing to the difference between the primarymagnetomotive force and the secondary magnetomotive
force. It was suggested that offsetting both themagnetomotive forces was incomplete in type-A.
Furthermore, the leakage magnetic flux due to theoffset magnetomotive force was considered to be a causeof an increase in eddy current loss in the winding.
Therefore, it was inferred that type-A had larger
winding loss at the large offset magnetomotive force
when the heavy load, compared with the other two typesof transformers. It was considered that the difference
between the losses of type-B and type-C was not only
due to the winding resistance of the secondary winding
but also due to the influence of the eddy current loss
caused by the leakage magnetic flux.

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017

137

95

10

Efficiency  [%]

90
85
1
80
75
70

Fig. 11 Circuit diagram of PFM controlled
quasi-resonant mode flyback-type DC-DC converter.
4. Application to PFM controlled quasi-resonant mode
flyback-type DC-DC converter
4.1 Evaluation circuit diagram and conditions
Since the PFM controlled quasi-resonant mode
flyback-type DC-DC converter7) used in this study
operates under the critical current mode and soft
switching operation, it is expected to realize both higher
conversion efficiency and higher switching frequency.
Fig. 11 shows the circuit diagram of the PFM
controlled quasi-resonant mode flyback-type DC-DC
converter with the composite magnetic core transformer
under investigation. While the drain-source voltage
waveform of the MOSFET was being confirmed with theoscilloscope, the switching frequency and on-time ratio
were adjusted by the function generator (YOKOGAWA;
FG420) so that the output voltage was constant at 12 V
even if the output current was changed.
4.2 Experimental results and discussion
Fig. 12 shows the measured power conversion

138: type-A: type-B: type-C0

10 20 30 40 50 60 70
Output power POUT [W]

Switching frequency fsw [MHz]type-A
type-B
type-C
Fig. 10 Magnetic flux density distributions of three types of transformers. Upper figure represents in-phase flux
component with exciting current, and lower figure shows flux component with 90-degree phase-lag from exciting
current when frequency was 1 MHz and load current was 10 A.

0.1

Fig. 12 Power conversion efficiency and switchingfrequency of converter using composite magnetic coretransformers with different winding layouts.
efficiency and switching frequency of the converter
using three types of composite magnetic coretransformers. In this evaluation condition, theswitching frequency varied from approximately 400 kHz
to 2 MHz, which was roughly in agreement with thefrequency band in which the power loss of the composite
magnetic core transformer was small in theabove-described operation analysis for the transformers.
As shown in Fig. 12, when type-C was used, themaximum power conversion efficiency was 92.5% at an
output power of around 28 W, and an efficiency of 90%
was obtained at an output power of 60 W.
Power conversion efficiency at the light load washighest when type-A was used. However, withincreasing the output power, the power conversion
efficiency decreased significantly. Main reason of theefficiency decrease at large output power condition wasconsidered to be owing to the increase of eddy current
loss in the winding, where the leakage flux generated by
the difference of the magnetomotive forces between

Journal of the Magnetics Society of Japan Vol.41, No.6, 2017

primary and secondary winding became large whenheavy load condition. On the other hand, in type-B andtype-C, the decrease in power conversion efficiency was
small even if the output power increases. These trends
were consistent with the results of power loss analysis
for the transformers.
5. Conclusion
In general, the low permeability magnetic core isnot suitable for transformer because of large leakage
inductance owing to the low coupling coefficient
between windings. Though the surface-oxidized
carbonyl-iron powder/epoxy composite magnetic core
has a low relative permeability of about 6, its magnetic
loss was small below 10 MHz in frequency.
This study was done to develop the surface-oxidized
carbonyl-iron powder/epoxy composite magnetic coretransformer with high coupling coefficient and to apply
it to the PFM controlled quasi-resonant mode
flyback-type DC-DC converter. The obtained results are
as follows.
(1) In order to obtain high coupling coefficient, theembedded winding structure was introduced to thecomposite magnetic core transformer design, and thethree kinds of winding layouts were investigated on thebases of 2D-FEA numerical simulation and experiments.
The adjacent layout of the primary and secondary
winding was best choice for highest coupling coefficient
over 0.99.
(2) The output voltage of the transformer, in which
the innermost of the primary winding and thesecondary winding were not adjacent to each other andhad low coupling coefficient, was higher than the value
calculated from the turn ratio at the no-load and lightload. This operation suggests that the operating point of
the transformer may unintentionally change from theoperating point designed based on the specifications of
the converter.
(3) The frequency at the minimum power loss
shifted to the low frequency side with increasing load
current in any transformer under investigation in thisstudy, and the transformers exhibited the minimum
power loss obtained at 500 kHz to 2 MHz according to
the load current intending for the PFM controlled
quasi-resonant type flyback converter in this study.
Since the switching frequency of the converter decreases
with increasing load current, there is a possibility thathigh efficiency can be maintained in the wide loadcondition when the composite magnetic coretransformer is applied to such converter.

(4) Application of the fabricated composite
magnetic core transformer with the best winding layout
described in above (1) to the 48 V input/12 V output
PFM controlled quasi-resonant mode flyback-type
DC-DC converter exhibited a maximum power
conversion efficiency of 92.5% at an output power ofaround 28 W, and 90% efficiency was obtained at an
output power of 60 W. The obtained efficiency
characteristics were consistent with the trend of power
loss analysis for the composite magnetic coretransformers.
Acknowledgements This study was supported in partby Grant-in-Aids for Japan Society for the Promotion of
Science (15H02233), New Energy and Industrial
Technology Development Organization (15101041-0),
Japan, and Japan Science and Technology Agency(Nagano Satellite Cluster, Kyoto Super Cluster
Program).
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Received June 13, 2017; Revised July 10, 2017; Accepted July24, 2017

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