Paper

Motor Performance Investigation of an Indirect
Matrix Converter with a Reactor Free Boost
Converter
Goh Teck Chiang* Student Member
Jun-ichi Itoh*
Member

Abstract— This paper describes a three-phase power converter, which utilizes the neutral point of an induction motor, anddemonstrates the motor performance of the proposed method under various operating conditions. Several motor test results, suchas the speed-torque characteristic, the torque impact and the acceleration-deceleration characteristic, are included to confirm theperformance of the induction motor. The structure of the proposed circuit consists primarily of an AC/AC converter (indirect
matrix converter (IMC)) and a boost converter connected to the neutral point of the induction motor. In addition, this paper also
proposes a method to reduce the ripple found in the battery current. Several experimental results showed that the motor works
well, and even the leakage inductance is utilized in the boost converter.
Keywords: Indirect Matrix Converter, Boost Converter, Induction Motor, Neutral point of the motor, Pulse Width Modulation

utilizing the leakage inductance of the motor at the connection of
the neutral point. In addition, the proposed circuit successfully
demonstrated the fundamental operation of good converter
performance.
However, the leakage inductance of a motor varies according to
the structure of the motor. This variation results if the battery
current consists of a ripple due to insufficient leakage inductance.
Therefore, more research is needed to investigate and confirm thestability and performance of the motor.
In this paper, the authors propose a simple pulse-widthmodulation (PWM) method that generates a high switchingfrequency to reduce the ripple in the battery without adding
additional hardware. The high-frequency switching method is only
applied to the secondary side of the proposed converter.
Furthermore, the performance of the connection of the neutral
point to the proposed circuit in a 750 W induction motor is also
examined in this paper. The stability and the performance of theinduction motor are investigated in an experiment. The
experimental results, which clarify the speed-torque curve, thetorque impact characteristic, and the acceleration-deceleration
characteristic, prove the validity of the proposed method.

1. INTRODUCTION
Hybrid electric vehicles (HEVs) are one of the promising fuelefficiency vehicles that are targeted to supply 20% of the total
number of vehicles in Japan by 2020. The research on HEVs has
proven this vehicle is able to reduce 48% of the CO2 emissions
released in traditional vehicles. Therefore, the known issues in
HEVs that are now actively studied are cost, size, battery technicalissues and high reliability of the electric motor.
The direct type of AC/AC converters has shown advantages inboth size and cost. Matrix converters formed by nine bidirectional
Reverse Blocking Insulated Gate Bipolar Transistors (RB-IGBTs)
can deliver output voltage at variable frequency without
electrolytic capacitors as storage devices [1]-[3]. However, thelack of a flexible connection to the DC link voltage in thisconverter is a problem for connecting a secondary source such asbatteries. In Ref. [4], it is shown that an extra inverter circuit isrequired for connecting the DC link voltage to the matrix
converter. Consequently, the size and cost would increase due tothe extra use of the components.
In contrast, the indirect (two-stage) matrix converter (IMC),
which also is a direct type of AC/AC converter constructed by acurrent source inverter (CSI) and a voltage source inverter (VSI),
has a stable DC link voltage that can be connected to an extra
source [5]-[7].
The authors previously proposed an AC/DC/AC circuit
applicable in the HEV system [8]. The proposed circuit greatly
reduced the size of the converter by removing two passive
components, the electrolytic capacitors and the boost reactor, by

2. CIRCUIT TOPOLOGY
Fig. 1 shows the IMC circuit structure with the DC link partconnected to a DC power source by a DC boost converter [8]. The
boost converter also connects to the neutral point of the motor andutilizes the leakage inductance of the motor as a boost reactor.
Fig. 2 illustrates the energy flow diagram of the proposed
circuit. In a hybrid system that has two power sources, a generator
and batteries, the proposed circuit is capable of operating in threemodes. The three modes are as follows: standard mode Pin= Pgen,* Nagaoka University of Technology
1603-1, Kamitomioka-machi, Nagaoka 940-2188
1

Primary
side

DC/DC
Converter

srp ssp stp

Secondary
side

ir *
is*
it*

scp sup svp swp
Motor

Lin

ir
is
it

200V
/50 HzCiniu
Edc

srn ssn stn

vo

iv
iw

DC linkripple cal

scn sun svn swn

+
-

Rec*

1/z

dq uvw vv*
conv.

100 V
ibat

ω*

Fig. 1 Proposed circuit topology.

V/f conv.

vq

vw*

*

PI
1
s

Fig. 4θ*

Pulse
patternconversion

Carrier
generator

vu*

vd*

0

10 kHzCarrierSingle
phase
Mod.

Ib* +

X
X
X
X
vc*

Sup

+
+ Svp
+ Swp
+
-

Scp

- Boost converter
Ib controller
Control block diagram.

The principle is to control the three-phase input voltage so that the
input voltage carrying the maximum phase voltage is always
selected; this selection corresponds to a switch that is always
turned on. Then, the other two legs consisting of the medium
phase voltage and the minimum phase voltage are in the switchingstate. The pulse pattern conversion is a simple logic selector thatconverts the voltage source PWM pulse pattern into the current
source PWM pulse pattern. For example, if Srp contains the
max-phase voltage, Srp = On, Ssp and Stp = Off at the upper arm,and Srn = Off at the lower arm. Here, Ssn and Stn are the switchingof the negative phase of the S phase and the T phase input voltages,
which are at the med-phase voltage and the min-phase voltage,
respectively. By applying this technique, a DC link voltage isformed that contains a frequency ripple with six times the input
frequency.
A carrier generator that can produce a variable carrier based on
the primary side command (Rec*) is used in the secondary sideand the boost converter side. The secondary side includes a V/fconversion to control the motor frequency and output voltage as
constant values [10]-[11]. The motor speed is adjusted
proportionally to the output frequency. Then, the magnetizing
current of the induction motor remains practically constant in awide adjustment range.
The boost converter controller contains a PI control for controlof the battery current, and then PI gain value is multiplied by theDC link frequency compensator to obtain the desired switchingpattern. In Ref. [8], it is explained that the boost converter
operates at every zero vector output of the inverter side.
3.2
Limitation of the Boost Converter Operation
Since the boost converter operating principle depends on the
inverter stage, the boost converter must apply the same type ofcarrier and the same frequency. One known limitation of thismethod is that the value of the inductance cannot be selected
because the leakage inductance corresponds to the motor structure.
Leakage inductance that is too small causes a high ripple in thebattery current. To overcome the high ripple, a higher switchingfrequency must be applied.
In addition, the application of this boost converter in the IMC
results in an inconstant switching duty ratio of the boost converter.
The secondary side of the IMC uses an inverter carrier that has afrequency that is two times higher than that of the original carrier.

Fig. 2 Proposed circuit energy flow diagram.

Fig. 3 Operation of the boost converter during the zero vectors.

electric vehicle (EV) mode Pin =Pbat, and hybrid electric vehicle(HEV) mode Pin = Pgen + Pbat. In these modes, Pin =input power,
Pbat = battery power, Pgen, = generator power.
The operation of the boost converter depends on the zero vector
periods of the inverter side. The battery current is charged throughthe motor leakage inductance at every zero vector period. Fig. 3
shows the equivalent circuit of the zero phase sequence. The upperarms of the inverter (Sup, Svp, Swp) are considered as one arm
during the zero vector periods. The output currents flow within the
inverter side, and the DC link current is zero. The primary sideachieves the zero current switching (ZCS) during this zero vector
period, as shown by the shaded switches in Fig. 3. The charge
mode operation of the boost converter is shown in Fig. 3. In the
charge mode, the battery current flows into the inverter arms andthen to the leakage inductance of the motor.

3. PROPOSED CONTROL METHOD
3.1 Conventional Control Method
Fig. 4 shows a control block diagram of the proposed circuit,which is based on the carrier comparison method. The primaryside uses single-phase modulation (Single phase Mod. in Fig. 4)
and includes a pulse pattern conversion to convert the voltage
source switching patterns into the current source [9], where ir*, is *and it* represent the input current commands. The conversion
technique of the single-phase modulation is explained in Ref. [9].
2

IEEJ Trans. EIS, Vol. 124, No.1, 2010

TITLETABLE I. SIMULATION PARAMETERS

10 kHz carrier
Rec*-1
1 Inverter carrier

10 kHz-1
Inverter1
zero vectors

12 kHz55 kHz0
Chopper upper arm (Scp)
10Battery current

1.6
0.8

T1
12.7

Ton

Toff

Ton Toff
Ton Toff
T2
12.8

200 V

Input frequency

50 Hz

Carrier frequency

10 KHzOutput frequency

35 Hz

DC source

100 V

Leakage Inductance

5 mHfrom one point to another. This phenomenon can be seen from thecarrier waveforms in Fig. 5. As a result, the battery current rippleincreases due to the inconstant primary command.
Fig. 5(a) and Fig. 5(b) show the simulation results of thebattery current ripple subjected to the inverter carrier. In Fig. 5(a),the primary command (Rec*) is located at the middle range of theoriginal carrier. The inverter carrier is formed with the same duty
per cycle, at approximately 20 kHz. The zero vector periods canbe considered constant in this range. From this figure, time T1 isnearly equal to T2. The chopper upper leg is switched on at every
20 kHz at the same time as that of the upper arm zero vector
periods. During the On-state (Ton), the battery current increases,
and it decreases during the Off-state (Toff).
In Fig. 5(b), Rec* is at the minimum range or the maximum
range of the original carrier (Rec* =0.00.2 or 0.81.0). In theseareas, the inverter carrier per cycling is not at the same frequency.
Hence, the zero vector periods are inconstant, as shown between
T1 and T2 of Fig. 5(b), and it can be stated that T1 is not equal to T2.
The battery current increases sharply during time T1. Until time T2,
the battery current cannot be released because the On-state andOff-state occur too rapidly. These periods cause the large ripple.
The following detail must be noted: an individual switchingfrequency that is different from the frequency of the inverter
carrier cannot be applied for the boost converter. If the switchingpattern of the boost converter fails to synchronize with the zero
vectors, the ripple is larger due to the incorrect switching timing.
3.3 Proposed Carrier Comparison Method
Fig. 6 shows the proposed control block diagram foreliminating the ripple in the battery current. If we only increase
the inverter frequency, the switching loss increases. Therefore, a
new inverter carrier that includes a limiter is introduced toeliminate the current ripple in the selected zone only. The
unselected zone remains the same as that of the conventional
carrier.
Compared with the conventional control diagram, all parts inthe proposed control diagram except the carrier controller are thesame. The inverter carrier is combined with carrier generator 1
and carrier generator 2 to output a new inverter carrier. Carrier
generator 1 carries two times the frequency of the original carrier,which is the inverter carrier from the conventional method, as
described in Section 3.2. Carrier generator 2 can double thefrequency of carrier generator 1. Carrier generator 2 uses a fixed
point duty command so that at every cycle an approximately
two-times higher frequency is generated. The symbol “2x” in Fig.
6 represents this multiplier. Then, these two carriers are sent to theblock controlled by the selector signal.
The selector decides whether carrier 1 or carrier 2 is the output
carrier for the boost converter controller and the secondary side
controller. The comparator, based on the comparison between theselector and Rec*, is used to distinguish the zone. That is, whenRec*= 0–0.25 or Rec*= 0.75–1.0, the comparator outputs signal(a) Rec*=0.5.
1

Input voltage

Time (ms) 12.9(b) Rec*= 0.2
Fig. 5 Increase of the battery current ripple due to the inconstant primarycommand Rec*.

Fig. 6 Modification of the carrier controller in the control block
diagram.

Fig. 7 Suppression of the battery current ripple with the proposed control.
(Rec*=0.2)

The offset value of the inverter carrier is adjusted according to thecross point of the primary side command and the normal carrier.
With this method, the switching periods for the zero vectors vary3

Fig. 8 Diagram explanation for the proposed control method.

“1” and carrier 1 is applied in this zone; thus, this zone is known
as the selected zone. However, when the unselected zone is within
Rec*= 0.26–0.74, the comparator outputs signal “0” with carrier 1.
Fig. 7 shows the switching pattern for the selected zone. This
figure uses the same period and condition shown in Fig. 5(b). It
can be seen that at time T1=T2, the battery current is well undercontrol. The zero vector periods are occurring constantly and thebattery ripple can be greatly reduced.
Fig. 8 explains the characteristics of the new inverter carrier. At
the top is the original 10-kHz carrier and Rec*. The second patternis the comparator signal. The third is that of the inverter carrier
from the conventional method (20 kHz). The last is the pattern ofthe proposed inverter carrier.
Each cycle of Rec* is separated into the selected zone and theunselected zone, as previously discussed. The selected zones are
the low zone within 0.0–0.25 and the high zone within 0.75–1.00.
When the comparator is “1” in the selected zone, a new carrier
having a double inverter frequency is used (40 kHz). When thecomparator is “0” in the selected zone, the same frequency of theoriginal carrier is copied into the new carrier. These frequencies
are used because these areas are too narrow to create a new carrier.
The unselected zone is applied with the conventional carrier,and the same frequency carrier is brought into the new carrier. As
a result, a new inverter carrier containing 20 kHz and 40 kHz
frequencies is formed. When the inverter carrier reaches theinconstant zone, the 40-kHz constant carrier takes over. Thismethod ensures the constant switching of the zero vectors for theselected zone.

4.

Fig. 9 Simulation result with the conventional carrier.

SIMULATION RESULTS

Table I shows the simulation parameters. Fig. 9 shows thesimulation results using the conventional type of inverter carrier.
The battery current contains a large ripple and also affects theinverter currents. It is known from Ref. [8] that the inverter
current contains one-third of the battery current as a zero-phasesequence component. Therefore, the total harmonics distortion(THD) value of the output current is influenced by the battery
ripple.
Fig. 10 shows the simulation results using the proposed method.
Note that the ripple of the battery current is greatly reduced. The
selectors are set in the range of 0.0–0.25 and 0.75–1.00 in thissimulation. The zone declaration can be clearly seen from theinverter carrier. The darker side represents the 40-kHz carrier andthe lighter side represents the 20-kHz carrier. The new inverter did
not affect the performance of the system and yet further reduced

Fig. 10 Simulation result with the proposed carrier.

Fig. 11 Harmonic analysis for the output current (iu).

the THD value of the output currents.
Fig. 11 shows the harmonics analysis for the output current (iu).
The top pattern is the conventional method and the bottom one is
4

IEEJ Trans. EIS, Vol. 124, No.1, 2010

TITLE

Fig. 12 Motor experimental setup.
TABLE II. MOTOR PARAMETERS

Induction motor (Fuji: MLH6085M)
Motor Power

750 W

Poles

4/ 50 Hz

Rated Voltage

200 V

Rated RPM

1420

3.6 A

Leakageinductance

4.42 mH

Rated Current

ES Servo motor (Fuji: GRK1151A)
Motor Power(a) Rec*=0.15-0.25.

1500 W

Rated Votlage

200 V

Rated Current

8.6 A

Rated RPM
Rated
TorqueMax Torque2000
7.16 N.m
10.8 N.m

TABLE III. EXPERIMENTAL PARAMETERSLC Filter Cut-off
frequency

2.4 kHz

50 Hz

DC source

100 V

Carrier frequency

10 kHzMeasured leakage
inductance

4.42 mH

Variable carrier
frequency

20 kHzOutput frequency

10 Hz –
45 Hz

Input voltage

200 V

Input frequency

Table III shows the experimental parameters. The verifications
are classified into three sections: the speed and torque curve, thetorque impact characteristic, and the acceleration and deceleration
characteristic.
5.1 Ripple Reduction in the Battery Current
Fig. 13(a) and Fig. 13(b) show the experimental results in thecase that the primary command (Rec*) is 0.15–0.25 and 0.75–0.85,
respectively. In these two figures, the top result shows the primarycommand pulse according to Rec*, the second shows the zero
vector pulse, the third is the switching pulse of the chopper upper
arm, and the last is the battery current. In Fig. 13(a), Rec* islocated at the low peak of the inverter carrier. The selected zone isset to start at Rec*=0.20, which is represented by the dotted line in
the middle of the figure. As can be seen from this figure, thebattery current contains a high ripple at the unselected zone. The
inverter carrier changes immediately once Rec* reaches 0.20, andthe zero vector periods occur rapidly. A1 and A2 in Fig. 13(a)show the periods where the carrier is larger than Rec*. B1 and B2
in Fig. 13(b) show the periods where the carrier is smaller thanRec*. Comparing periods A1 and A2, the zero vectors in period
A2 are approximately twice those in period A1 due to the use of
the 40-kHz carrier. Notice that the conventional 20-kHz carrier isapplied in periods B1 and B2. As a result, the amplitude of theripple is reduced.
Fig. 13(b) shows the result in the case that Rec* is located at thepeak of the inverter carrier. The frequency of the inverter carrier(b) Rec*=0.75-0.85.
Fig. 13 Relations between the zero vector pulses and the battery current
ripple with the proposed method.

the proposed method. At 10 kHz, the 2nd harmonic component of
the proposed method is less than that of the conventional method.
This result indicates that the THD of the output current is
improved.

5. EXPERIMENTAL RESULTS
The first subsection of this section discusses the experimental
results of the ripple reduction in the battery current, the second
section shows the adjustable speed drive performances with thethree-phase induction motor used in the proposed circuit, the third
shows
the
torqueimpact,andthefourththeacceleration-deceleration characteristic. The motor performances
are investigated based on the control block diagram in Fig. 4.
Fig. 12 shows the setup between the induction motor and theload generator, which uses a servo motor. Table II shows theparameters of the induction motor and the servo motor. The servo
motor provides the motor load. The torque sensor and the speed
sensor are located at the servo motor. The induction motor is
powered by the indirect matrix converter, and the servo motor is
directly connected to the grid. A controller is attached to the servo
motor to control the torque directly. The speed and torque data are
obtained from the controller for monitoring purposes.
5

120
100

20 Hz 25 Hz 30 Hz 35 Hz
15 Hz

Torque (%)

80
60

10 Hz

40 Hz
45 Hz

40
20
0

0

200

400

600
800
Speed (rpm)

1000

1200

1400

(a) Indirect matrix converter.

Torque (%)(a) Indirect matrix converter.

(b) Proposed converter.
Fig.14 Speed and torque characteristic of the induction motor.
(b) Proposed circuit in EV mode. (Pbat =200 W)
Fig. 15 Torque impact characteristic of the induction motor.

changes to 40 kHz immediately after Rec* is 0.80. In theunselected zone (Rec* < 0.80), the amplitude of the ripple is highdue to the inconstant switching period. However, in the selectedzone (Rec* > 0.80), the ripple is reduced because of the highswitching frequency and constant switching duty ratio. The
definitions of A1, A2, B1 and B2 are the same as those for Fig.
13(a). Notice that the sequence of the zero vector in period B2
increases more than that in period B1. In both (a) and (b), theripple is approximately reduced by 42%.
5.2 Speed and Torque Curve
Fig. 14 shows the relationships between the speed and torquecharacteristics of the induction motor. Fig. 14(a) shows the
motoring data in the case that only the IMC is connected. Notethat the V/f control is applied at all times. This graph shows good
control over the induction motor. The reason the speed is notconstant to the torque is because the induction motor contains slip
according to the load torque.
Fig. 14(b) shows the relationships between the speed andtorque characteristics of the proposed circuit connected to the
induction motor. The battery current is pre-set at 0 A when thetorque is below 20%. When the torque is 20% or larger, the battery
current increases to 2 A. At an output frequency of 35 Hz, thetorque is 100%, and the motor speed is 879 rpm. Likewise for the
IMC connection under the same output frequency, the torque is100%, and the motor speed is 877 rpm. The data in these twographs show no difference, even at the neutral point of the motor
connected to the boost converter. That is, the motor with theproposed converter runs in good operating condition.
Note that at 40 Hz and 45 Hz, the torque cannot reach 100%
because of the limitation of the output voltage transfer ratio of the
IMC. The output voltage transfer ratio of the IMC is limited to
0.866 of the input voltage. Moreover, the proposed circuit must

apply three-phase modulation in the secondary side in order togenerate the zero vector periods. This further reduces the voltage
transfer ratio from 0.866 to 0.75. Therefore, starting at the output
frequency of 37.5 Hz, the output voltage is not sufficient to deliver
the power response to the torque.
5.3 Torque Impact Characteristic
This section demonstrates the torque impact characteristic
under the EV mode and the normal mode, as shown in Fig. 15. Fig.
15(a) shows the normal mode operation, in which the boost
converter is not connected to the motor and only the IMC isoperating. The output frequency is 30 Hz, and the step increase of
the torque is 100%
In Fig. 15(a), input current ir shows an unbalanced sinusoidal
waveform due to the low input power, which is nearly 170 W. As
the torque increases to 100%, the input power immediately
produces more power to maintain the speed. The output current iu
shows a good sinusoidal waveform at all times
The performance of the motor under the EV mode is shown in
Fig. 15(b). The experimental condition is the same as that in Fig.
15(a). The battery power is 200 W at all times, and the input
power is nearly 50 W; therefore, ir shows poor waveform quality
due to the low input current. When the torque increases to 100%,
the battery power is maintained at 200 W and the input power
eventually increases to maintain the motor speed. The input
current ir shows an unbalanced phenomenon due to the low input
power, because the total input power is partially supplied by thesecondary power. The output current iu demonstrates a goodsinusoidal waveform at all times, which matches the result in Fig.
15(a).
5.4 Acceleration and Deceleration Characteristic
6

IEEJ Trans. EIS, Vol. 124, No.1, 2010

TITLE

25

Pout =800W

Filter loss

Loss [W]20

Recovery
loss

15

70 %

95%

Switchingloss

10

Conduction
loss

5

87%

40%

(a) Indirect matrix converter.
0PS(a) 20 kHz

B

P

S

B

(b) Proposed methodPS(c) 40 kHz

B

Fig. 17 Loss comparison among conventional methods and proposed
method.

the three results. Comparing (a) and (b), the switching loss isincreased by 70%. However, comparing (a) and (c), the switchingloss is increased by 95%, which is higher than that of the proposed
method. As for the boost converter, comparing (a) and (b) shows
approximately a 40% increase, but comparing (a) and (c) shows
approximately a 87% increase. Overall, the total loss for (a) is 31
W, (b) is 37 W, and (c) is 41 W. The proposed method is able toachieve high efficiency and also to reduce the ripple in the battery
current.
Studies on motor losses were also investigated by the authors.
(See Refs. [13]-[14].) In the present proposed method, the battery
current is added to the output current; therefore, the copper loss isincreased proportionally. However, the copper loss in the rotorside remains unchanged because the zero-phase current (battery
current) does not produce flux to rotate the rotor. Refs. [13] and[14] are studies of an internal permanent magnet (IPM) motor,
where the copper loss is increased by 26% compared to that of the
proposed method and the conventional method. The copper loss of
the induction motor is also considered to increase. The studies
show a trade-off relationship of the efficiency between theconverter and the motor.

(b) Proposed converter.
Fig.16 Acceleration and deceleration characteristic.

This section demonstrates the speed characteristic of theinduction motor under two different setups, normal mode andHEV mode. This experiment was conducted under a no-loadcondition. For the acceleration, the speed command increases from300 r/min to 900 r/min. For the deceleration, the speed commanddecreases from 900 r/min to 300 r/min.
Fig. 16(a) shows the IMC only (normal mode) data. During
acceleration, large input power is required until the speed isconstant. The output current iu shows a good sinusoidal waveform
and the change of the shape of the frequency is due to theacceleration and deceleration control.
Fig. 16(b) shows the proposed circuit speed characteristic.
When the motor speed is low (300 r/min), the battery current is set
at 0 A. Once the acceleration starts, the battery power supplies 200
W immediately. Therefore, during acceleration the input current ir
has a low magnitude because the battery power is used to drive themotor instead providing the input power. When the speed isconstant, the battery power continues delivering and the input
power drops to the minimum.
Comparing Fig. 16(a) and Fig. 16(b), it can be seen that theoutput current iu shows a perfect sinusoidal waveform. That is, theconnection of the neutral point of the motor for using the boost
converter does not affect the motor performance. Furthermore, theresults demonstrate good control of the power management with
the input and the battery.
5.5 Loss Analysis and Motor Efficiency
Fig. 17 shows the comparison of the losses among theconventional methods and the proposed method. The simulation
parameters are the same as those listed in Table I. The battery
current is controlled to 1 A. To clarify the advantage of the
proposed method (b), the conventional method is operated at two
frequencies: (a) 20 kHz and (c) 40 kHz. In this figure, P is theprimary side, S is the secondary side and B is the boost converter.
The conduction loss for the secondary side remains unchanged in

6. Conclusion
A simple high-frequency PWM method is proposed to reduce
the ripple in the battery current. The simulation and experimental
results confirm the validity of the proposed method. In addition,
this paper discusses the performance of an induction motor, inwhich the neutral point of the motor is connected to a DC boost
converter. Several experimental results showed that the motor
works well, and even the stator leakage inductance is utilized in
the boost converter.
(Manuscript received -. -, 2010, revised - -, 2010)
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(2)

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using an Indirect Matrix Converter with an Active Snubber Circuit”, 13th
Power Electronics and Motion Control Conference, Poznan, pp.581-588,
Sept. 2008(13 ) Itoh, J-i., Ikarashi, D.: “Investigation of a two-stage boost converter using
the neutral point of a motor”, International Power Electronics Conference(IPEC), Sapporo, pp. 1180, 21-24 June 2010(14) Itoh, J-i., Ikarashi, D.: “Loss evaluation of a two-stage boost converter
using the neutral point of a motor”, IEEE Energy Conversion Congress andExposition (ECCE), Atlanta, GA, pp. 2600, 12-16 Sept. 2010

(Student member) received his B.S. degree inelectrical and electronic engineering from
Queensland University of Technology, Brisbane,
Australia in 2004 and his M.S. degree in electricaland electronic engineering from Nagaoka Universityof Technology, Japan in 2009, where he is currently
working towards to a Ph.D. degree in electrical andelectronic engineering. His research interests include
indirect matrix converters and PWM.
Goh Teck Chiang(Member) received his M.S. and PhD degrees inelectrical and electronic systems engineering from
Nagaoka University of Technology, Nagaoka, Japan
in 1996 and 2000, respectively. From 1996 to 2004,
he was with Fuji Electric Corporate Research andDevelopment Ltd., Tokyo, Japan. Since 2004, He
has been with Nagaoka University of Technology as
an associate professor. He received the IEEJ
Academic Promotion Award (IEEJ Technical Development Award) in 2007.
His research interests include matrix converters, DC/DC converters, power
factor correction techniques and motor drives. He is a member of theInstitute of Electrical Engineers of Japan.
Jun-ichi Itoh8

IEEJ Trans. EIS, Vol. 124, No.1, 2010