Zero-Voltage Switching for Bidirectional Buck/Boost
Converter using Hybrid Discontinuous Current Mode
Hoai Nam Le, Koji Orikawa, Jun-ichi Itoh
Nagaoka University of Technology
1603-1 Kamitomioka Town
Nagaoka City, Niigata Prefecture, Japan
Tel.: +81 / (258) – 47.9533.
E-Mail: lehoainam@stn.nagaokaut.ac.jp
URL: http://itohserver01.nagaokaut.ac.jp/itohlab/index.html
Abstract— This paper proposes a hybrid current mode between Triangular current mode (TCM) and Discontinuous
current mode (DCM) in order to achieve Zero-voltage switching (ZVS) for a bidirectional buck/boost DC-DC converter. In
the proposed Hybrid Discontinuous Current Mode entitled
HDCM, the TCM operation is applied during the zero-current
interval of DCM. Therefore, both ZVS and the variable current ripple, which result in the high efficiency at wide load
range, are achieved. The achievement of ZVS with HDCM isconfirmed by a 600-W prototype. Compared to TCM, theRoot-Mean-Square current is reduced by 47.2% at most with
HDCM, which further contributes to the loss reduction. Moreover, under the condition of the same boost inductor, the efficiency of HDCM at load of 0.2 p.u. is improved by 1.5% compared to TCM.
Keywords— Zero-voltage switching; High power density;
Discontinuous current mode; Triangular current mode

I. INTRODUCTION
Over the past decades, the increase of the power density
has been one of the most important design criteria in powerelectronics [1]. The minimization of the converters provides
not only the material cost reduction but also the easy implementation for applications which require space-saving powersystems such as, e.g. Hybrid electric vehicles or Photovoltaic
systems. The boost inductor is one of the components which
contributes mainly to the converter volume. By increasing
the switching frequency, the volume of the boost inductor
can be reduced. However, higher switching frequency leads
to the increase of the switching loss, which requires larger
heat sinks. Therefore, the reduction of the switching loss is
crucial to the high power density design.
The soft switching technique can reduce greatly theswitching loss, in which the resonance between the inductor
and the capacitor is utilized in order to achieve the Zerovoltage switching (ZVS). However, in order to satisfy thecondition for ZVS, these methods suffer many drawbacks;
e.g. the requirement of the additional components [2], or the
restriction of the controllable duty ratio [3]-[4]. Consequently, the additional components not only restricts the minimization of the converter but also complicates the control

978-1-5090-1815-4/16/$31.00 ©2016 IEEE

method, whereas the restriction of the voltage ratio or the
duty ratio limits the application of ZVS. Therefore, theachievement of ZVS without additional components or thelimited duty ratio is desired.
On the other hand, ZVS can be achieved with an advanced modulation technique, entitled Triangular current
mode (TCM) [5]-[7]. In particular, in Continuous current
mode (CCM), the free-wheeling diodes suffer from significant reverse recovery losses. The reverse recovery issues canbe alleviated by operating the converter in Discontinuous
current mode (DCM). Nevertheless, in DCM, it is still necessary to deal with two kinds of switching loss; the high turnoff loss because the switching devices are turned off at least
twice of the average current, and the turn-on loss due to thejunction capacitor charge. In TCM, the turn-off loss can be
decreased greatly by connecting a snubber capacitor in parallel with the switching devices, whereas the turn-on loss iseliminated by ZVS. As a result, the switching frequency canbe pushed to several MHz due to almost no switching loss.
One of the main drawbacks in the TCM operation is that thehigh current ripple is constant at all load range. As a result, it
is difficult to achieve high efficiency at light load when the
average current is low [5]. Therefore, it is desired to reducethe current ripple at light load in order to avoid the sharp
decrease in the efficiency.
This paper proposes a novel concept of the hybrid current
mode between TCM and DCM, which is entitled Hybrid
Discontinuous current mode (HDCM), in order to achieveboth ZVS and the variable current ripple. In the proposedcurrent mode, the TCM operation is applied during the zerocurrent interval of the DCM current in order to generate thecondition for ZVS. As a result, this current mode can achieve
the high efficiency at wide load range without additional
components. This paper is organized as follows; first, themechanism of the ZVS achievement in HDCM is described
and the control method of HDCM is explained. Next, theprinciple operation of HDCM is confirmed by experiments
and the comparison of the Root-Mean-Square (RMS) current
between TCM and HDCM is conducted. Finally, the efficiency between TCM and HDCM is compared in order toconfirm the effectiveness of HDCM.

II. HYBRID CURRENT MODE BETWEEN DISCONTINUOUS
CURRENT MODE AND TRIANGLE CURRENT MODE
Fig. 1 depicts the bidirectional buck/boost converter. In
order to minimize the passive components, i.e. the boost inductor L and the output capacitor C, the switching frequency
is required to be pushed to hundreds of kHz or even to several MHz. Hence, the switching loss reduction is crucial to thehigh power density design. There are three conventional current modes to operate the bidirectional buck/boost converter:
CCM, DCM and TCM. Note that the Critical Current Mode
(CRM) operation can be analyzed as same as the DCM operation without the occurrence of the zero-current period.
Fig. 2 illustrates the pattern of the switching losses ineach current mode. In CCM shown in Fig. 2(a), the significant reverse recovery loss occurs at the beginning of the interval 1, because the free-wheeling diode FWD2 is forcefully
turned off. In DCM shown in Fig. 2(b), due to the natural
turn-off of the free-wheeling diode FWD2 at the end of theinterval 4, the reverse recovery loss does not occur. Nevertheless, the junction capacitor discharges through the switch
FET1 at the beginning of the interval 1 in Fig. 2(b), which
results in the hard switching. On the other hand, in TCM
shown in Fig. 2(c), the current commutates from the switch
FET2 to the free-wheeling diode FWD1 during the interval 4,
and gradually discharges the junction capacitor. Hence, theswitch FET1 is turned on at the forward voltage of the freewheeling diode FWD1, which is considered as ZVS. As aconclusion, the principle of the ZVS achievement in TCM is
to flow a negative current in order to discharge the junction
capacitor. Besides, it should be noted that the turn-off loss
can be reduced simply by connecting a snubber capacitor inparallel with the switch [7]. However, a drawback of TCM isthe large current ripple which decreases notably the efficiency at light load. Therefore, the current ripple reduction of
TCM at light load is desired.
In order to reduce the current ripple, one of the conventional methods is to increase the switching frequency at light
load, i.e. the Pulse Frequency Modulation (PFM) [6]. However, PFM is undesirable in many power electronic systems
because it is difficult to design a filter circuit for the operation across a wide range of frequencies. On the other hand, inthe DCM operation, because the current ripple becomes
smaller at light load, the high efficiency can be maintained.
As mentioned above, because the DCM operation still suffers the high turn-on loss, the method to achieve ZVS in
DCM is desired. Therefore, this paper proposes the novel
concept where TCM and DCM are combined in order toutilize the advantage of both TCM and DCM, i.e. ZVS in
TCM and the current ripple reduction at light load in DCM.
Fig. 3 illustrates the concept of the hybrid current mode
(HDCM) between DCM and TCM. During the zero-current
interval in DCM, i.e. the interval 4 and 5 in Fig. 2(b), instead of let the current return to zero, the switches are
modulated in order to flow the TCM current. As a result, thecondition in order to achieve ZVS in the DCM operation
can be satisfied by the TCM current during the interval 4 inFig. 3. Note that the switching losses during the TCM operation are negligibly small as explained above. Further-more, because the total charge during the TCM interval can
FET2
L

SW2

FET1

iL

A

SW1

Vin

Cds2

FWD2

Load
Vout

C
FWD1

Cds1

Fig. 1. Bidirectional buck/boost DC-DC converter. The switching loss
reduction is crucial to the high power density design. In general, there arethree current modes to operate the converter: CCM, DCM and TCM.
Gate Signal

Tsw

Deadtime

Deadtime

FET2ONFET1ONFET1ONt0iL

Psw_off

ZVS

Psw_recovery

ZVS

t

0
Mode:12

3

41(a) Switching losses in CCM. In the CCM operation, the free-wheeling
diodes suffer from significant reverse recovery losses.
Gate Signal
FET1ON

Tsw

Deadtime
FET2ONFET1ONt0iL

Psw_off
Psw_on

ZCS

t

ZVS

0
Mode:1

2

3

4

5

1

(b) Switching losses in DCM. The reverse recovery issues can be alleviated by DCM. Nevertheless, in the DCM operation, two kinds of switching loss still occur; the high turn-off loss because the switching devices isturned off at least twice of the average current, and the turn-on loss due tothe junction capacitor charge of the free-wheeling diodes.
Gate Signal

Tsw

Deadtime
FET1ON

Deadtime

FET2ONFET1ONt0iL

Psw_off

ZVS
t0

Psw_off
ZVS
Mode: 1

2

3

4

1

(c) Switching losses in TCM.
Fig. 2. Switching losses in CCM, DCM and TCM. In the TCM operation,
the turn-off loss can be decreased greatly by connecting a snubber capacitor in parallel with the switching devices, whereas the turn-on loss iseliminated by ZVS by using the inductor to resonate out the junction capacitor charge [6]. By utilizing TCM, ZVS is achieved without additional
components or the limited duty ratio. However, the high current ripple isconstant at all load range, which results in the shape decline of the efficiency at light load.

be controlled to become negligibly small, the TCM interval
is considered to have no effects on the DCM operation. On
the other words, the current ripple at light load can be reduced by the DCM operation. Consequently, both ZVS andthe current ripple reduction at light load are achieved without additional components or the limited duty ratio.
III. CONTROL OF HYRID DISCONTINOUS CURRENT MODE
Because HDCM is combined of two current modes, i.e.
DCM and TCM, the DCM feedback current control and the
TCM switching pulse generation are introduced. In particular, the DCM nonlinearity which occurs in the duty-ratio-tocurrent transfer function due to the zero-current interval,
worsens the current control performance if the same PI controller is applied as in CCM [8]-[10]. Therefore, in this section, first, the current feedback control with the DCM nonlinearity compensation is introduced. Next, the generation of
the duty cycles for the TCM operation during the zerocurrent interval is explained.
A. Discontinous current mode feedback current control
Fig. 4 depicts the inductor current waveform in DCM,
where D1, D2 and D0 denote the duty ratios of the first, thesecond and the zero-current interval. Note that the output
voltage Vout is assumed to be constant in the design step of
the feedback current control because the inductor current
response is generally designed to be much faster than the
output voltage response. The equation based on the averagemodel of the boost/buck converter in DCM is given by (1),diL _ avg

 D1Vin  D2 Vin  Vout  ........................... (1)dt
where Vin is the input voltage. The average current iL_avg andthe current peak iL_peak, which are shown in Fig. 4 are expressed as,
LiL _ avg 

iL _ peak2( D1  D2 ) ...................................... (2)Vin
D1Tsw ................................................. (3)L
where Tsw is the switching period. Substituting (3) into (2)and then solving the equation for the duty ratio D2. The duty
ratio D2 is expressed by (4),2 LiL _ avg
D2  D1 ............................................... (4)D1TswViniL _ peak 

Substituting (4) into (1) in order to remove the duty ratio D2
and represent (1) as a function of only the duty ratio D1,then (5) is obtained [8],diL _ avg
2 LiL _ avg
L Vin  Vout  D1Vout  (Vout  Vin )(1 )
dt
Vin D1Tsw.............................................................................. (5)Next the duty-ratio-to-current transfer function in DCM
Gi_DCM is obtained by linearizing (5) at the steady-state
operating points [9],iL _ avg ( s)2Vdc
Gi _ DCM ( s) 2(Vout  Vin ) L ....... (6)D1 ( s)sL Vin D1 _ sTsw
where iL_avg and D1 are the small signals of the average
current iL_avg and the duty ratio D1, and D1_s is the duty ratioof SW1 at the steady-state operating points. On the other
hand, the duty-ratio-to-current transfer function in CCM
was derived in [9], yielding
iL _ avg ( s) Vdc
Gi _ CCM ( s) ................................ (7)D1 ( s)sL
Fig. 5 shows the gain of the duty-ratio-to-current transfer
function in CCM and DCM under different conditions of thesteady-state duty-ratio D1_s. In most cases, the frequency
corresponding to the pole of Gi_DCM is certainly much higher
than the cutoff frequency of the current control loop fn. Consequently, the open loop gain in DCM is much lower than in
CCM. This worsens the current response in DCM if the same
PI controller as in CCM is employed in DCM. Therefore, theoutput of PI controller is necessary to be compensated when
the circuit is operated in DCM in order to achieve the same
current command response as in CCM.
Fig. 6 illustrates the circuit model of the buck converterin the DCM operation which is based on (6). In CCM, thedash line part does not exist, because the average current
iL_avg equals to the half current peak iL_peak/2. On the otherwords, this makes the zero-current interval D0Tsw in Fig. 5
become zero. However, in DCM, the voltage at point A in
Fig. 4 occurring during the zero-current interval introduces
the nonlinearity into the DCM transfer function. In order todesign of the compensation part for the nonlinearity of DCM,
the circuit model in Fig. 7 is linearized at the steady-state
operating points.

160

Vin

Vout

140
CCM

120

Gain [dB]

100
80
60
40

DCM

Boost Inductance L: 74 mH
Input Voltage Vin: 150 V
Output Voltage Vout: 285 V
Switching Period Tsw: 10 ms

D1

Do not exist in
CCM

D3
1
iL_avg0.5iL_peak

VinTsw
2L 0.5iL_peak

0
D1_s = 0.05

-20

iL_avgVout-Vin

D1_s = 0.3

20

-40
0.1

1
sL

Vout

iL_avgNonlinear factor
1

10

100
1k
10k 100k 1M 10M
Frequency f [Hz]Fig. 5. Duty-ratio-to-current transfer function for CCM and DCM. The
open loop gain in DCM is much lower than in CCM. This worsens thecurrent response in DCM if the same PI controller as in CCM is employedin DCM [9]-[10].

Fig. 6. Circuit model of boost/buck converter in DCM operation. The
zero-current interval D0Tsw makes the duty-to-current transfer function
nonlinear and reduces the open loop gain in DCM.
Designed as in CCM
iL_avg*Fig. 7 depicts the proposed DCM feedback current control block [11]. Note that the PI controllers in the proposed
method are designed as in CCM. The principle of the DCM
nonlinearity compensation is the estimation of the duty ratioat the steady-state operating points in order to compensate
for the nonlinearity in DCM. In the proposed method, theduty ratio at the stable points is estimated by utilizing theduty ratio at the previous calculation period. This provides
the control system circuit-parameter-independence, and short
calculation time. Moreover, because the dash-line part in Fig.
7 always compensates for the nonlinear factor in Fig. 6 byutilizing the duty ratio at the previous calculation period, thenonlinearity of DCM is eliminated in entire load range. This
enables the stability of DCM to be designed as same as oneof CCM, which has been designed and analyzed widely.
B. Switching signal generation for triangular current mode
Fig. 8 depicts the flowchart of switching signal generation during TCM interval. In HDCM, it is necessary to control the turn-on period of the switches during the TCM interval as shown in Fig. 3 in order to satisfy the following conditions;(i) the TCM interval has no influence on the DCM operation, i.e. the total charge during the TCM interval is required
to be negligibly small,(ii) the ZVS condition for the next DCM period is satisfied, i.e. the current value at the end of the TCM interval isrequired to be adjusted according to the zero-current intervalof DCM by controlling the TCM current peak and the TCM
period,(iii) the losses from the TCM interval such as the conduction loss are minimized, i.e. the current peak during the TCM
interval is large just enough to achieve ZVS for the TCM
operation.

aPI

iL_avg

a/b

D1

Vin
Vout-Vinz-1b

Circuit modelin Fig. 6

Nonlinearity Compensation
Control System Circuit

Fig. 7. Proposed DCM feedback current control block [11]. The principleof the proposed method is that, the nonlinearity compensation isconstructed by utilizing the duty ratio at the previous calculation period.
Start
Input parameters: Vin, Vout, Cds, Tsw, D1
Cal. of min. required current peak: Ipk_min, Eq. (8)Cal. of min. required TCM period: TTCM_min, Eq. (9)Cal. of zero current period: Tzero, Eq. (10)Cal. of TCM period number: nTCM, Eq. (11)Cal. of applied TCM period: TTCM , Eq. (12)Cal. of applied current peak: Ipk, Eq. (13)Generating switching pulses: TR1, TR2, Eq. (14)-(15)EndFig. 8. Flowchart of switching signal generation during TCM interval.
Because the TCM current is flown in order to achieve the ZVS condition
for the DCM operation, the losses and the charge from the additional
TCM current is required to be minimized. Furthermore, ZVS is also required to be achieved during the TCM interval.

Table 1 Experimental Parameters.
150 V
Vin
Vout 285 V
600 W
Pn
Rated Power
100 kHzf
Switching Frequency
sw
Td
200 nsDead TimeL
74 mH
Boost Inductor
C
10 mF
Output Capacitor

Gate Signal
FET2ONFET1ONFET1ONFET2ON

Input Voltage
Output Voltage

t

0
vCds
vCds2
Vout

vCds1

Switching Device: TPH3006PS

t

0
iL

TR2 Dead.

TR1

TR1

Dead. TR2TR2nTCM 

Ipk
t
0
IR

Qc Qc
Qc=Cds*Vout-Ipk
nTCM: 1

2

3

TTCM

TTCM

TTCM

3

Mode:
4

Tzero

TTCM _ min; nTCM is an odd number. .......... (11)Next, the actual TCM period TTCM is calculated from thezero-current period Tzero and the number of the TCM period
nTCM,

Tzero
....................................................... (12)nTCM
Then, the actual current peak Ipk of the TCM current is expressed as in (13),
TTCM 

1

Fig. 9. Flowchart of switching signal generation during TCM interval. In
order to achieve ZVS for the DCM operation, the TCM period TTCM isadjusted according to the zero-current period Tzero. Furthermore, in order tominimize the losses from the additional TCM current, the current peak Ipk
is designed to be just large enough to achieve ZVS for both switches SW1and SW2 during the TCM interval. Finally, in order to make the TCM
interval have no influence on the DCM operation, which implies the total
charge from the additional TCM current is negligibly small, the positive
and negative values of the current peak Ipk is controlled to be the same.

Fig. 9 illustrates the waveforms of the gate signal, thejunction capacitor voltage and the current during the TCM
interval in case of Vin > Vout/2. First, the minimum current
peak Ipk_min which achieves the condition of ZVS for SW1 iscalculated from the total charge stored Qc in the junction
capacitor Cds as in [6],yielding

I pk _ min Tzero

2C dsVoutVin....................................... (8)L

In order to let the inductor current reach the peak Ipk_min, therequired TCM period TTCM_min is calculated by (9),
1
1 
TTCM _ min  I pk _ min L  .................. (9) Vout  Vin Vin Next, the number of the TCM period nTCM is calculated fromthe zero-current period Tzero which is shown as D0Tsw in Fig.
4. The zero-current period Tzero is estimated from the duty
ratio of SW1 which is generated from the current control in
Fig. 7, yieldingVout 
 ............................... (10)Tzero  Tsw 1  D1
Vout  Vin In order to minimize the losses from the TCM interval, thecurrent peak Ipk of the TCM current is required to be minimal. Consequently, it is necessary to choose the number of
the TCM period nTCM as a maximum odd number, yielding

TTCM

1
1  ........................................ (13)L  Vout  Vin Vin Finally, the duty ratios TR1 and TR2of SW1 and SW2 in orderto achieve the TCM current peak Ipk is expressed as in (14)and (15), respectively [6],I pk 

TR1 

2C V VL2
I pk
 ds out in ............................... (14)Vin
L

TR 2 

2C V (V  V )L
2
I pk
 ds out out in .......... (15)
Vout  Vin
L

As shown in Fig. 9, the positive and negative values of thecurrent peak Ipk is controlled to be the same, which enables
the total charge from the additional TCM current to be negligibly small. Consequently, the TCM interval has almost no
influence on the DCM operation.
IV. EXPERIMENTAL RESULTS
Table 1 shows the experimental parameters. The designof the converter with high switching frequency and high inductor current in HDCM benefits the reduction of the boost
inductance. Note that the inductor is designed in order tooperate the converter in TCM at rated load. Besides, the design of the output voltage control is as same as one in the
CCM operation because the DCM nonlinearity is eliminatedcompletely.
Fig. 10 depicts the waveforms of the TCM and HDCM
current, the junction capacitor voltage, and the gate signal
under different conditions of load. At the rated load, the current waveforms are same for both HDCM and TCM. However, at light load, compared to the TCM operation, the current ripple is greatly reduced by HDCM. In particular, at thecondition of light load of 0.2 p.u., the RMS current is re-duced from 2.9 Arms to 1.5 Arms. Furthermore, as shown in
Fig. 10(d)-(e), the positive and negative values of the TCM
current are controlled to be almost the same. Consequently,
the addition TCM current has no effects on the DCM operation.

Consequently, almost no switching loss can be accomplishedin order to further increase the switching frequency andachieve the high power density. Note that there are no surge
voltage occurring in the junction voltage of both switches
SW1 and SW2 due to the achievement of soft switching.

Fig. 11 illustrates the waveforms of the gate signal andthe junction capacitor voltage, which is the magnified waveforms from Fig. 10(e). It is confirmed that both the switches
SW1 and SW2 are turned on at the zero voltage, which result
in ZVS. Furthermore, the turn-off losses can be greatly reduced by connecting a snubber capacitor parallel to theswitch in order to delay the increase of the junction voltage.

Fig. 12 illustrates the current response of HDCM. The
TCM current peak Ipk is controlled according to the zerocurrent period Tzero, which is dependent on the average current iL_avg. Consequently, ZVS for both switches SW1 andSW2 is achieved in all load range, i.e. the ZVS achievement
is unlimited by the condition of load or the duty ratio. Besides, there is a small oscillation occurring in the average

Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)

0V

0V

0A
Inductor Current iL (5 A/div) Time (4 ms/div) RMS current = 4.9 Arms(a) TCM and HDCM current and gate signals at rated load of 1.0 p.u..
Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)

Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)

0V

0V

0V

0V

0A

0A

Inductor Current iL (5 A/div) Time (4 ms/div) RMS current = 3.7 Arms Inductor Current iL (5 A/div) Time (4 ms/div) RMS current = 2.9 Arms(b) TCM current and gate signals at light load of 0.6 p.u..

(c) TCM current and gate signals at light load of 0.2 p.u..

Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)

Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)

0V

0V

0V

0V
11(a)

11(b)

0A

0A

Inductor Current iL (5 A/div) Time (4 ms/div) RMS current = 3.3 Arms

Inductor Current iL (5 A/div) Time (4 ms/div) RMS current = 1.5 Arms(d) HDCM current and gate signals at light load of 0.6 p.u..
(e) HDCM current and gate signals at light load of 0.2 p.u..
Fig. 10. Waveforms of TCM and HDCM current, junction capacitor voltage, and gate signal under different conditions of load. By utilizing the characteristic of
the zero-current interval in DCM, the current ripple of HDCM is reduced at light load. In particular, compared to the TCM operation, the RMS current in the
HDCM operation is reduced from 2.9 Arms to 1.5 Arms at light load of 0.2 p.u..

current iL_avg. This is because the sampling frequency as wellas the control frequency is set as 10 kHz which is lower 10times than the switching frequency of 100 kHz. Accordingly,
this problem can be solved simply by designing the control
frequency equal to the switching frequency.
Fig. 13 shows the comparison of RMS current and efficiency between TCM and HDCM. First, in the results of theratio between the RMS current iL_rms and the average current
iL_avg, by utilizing the variable current ripple of DCM, theRMS current in the HDCM operation is reduced greatlycompared to the TCM operation. In particular, at light loadof 0.2 p.u., the RMS current of HDCM is reduced by 47.2%.
Besides, the RMS current of HDCM at rated load is as sameas one in TCM, because the converter is designed to be operated with TCM at rated load. This design minimized the required current ripple in order to achieve ZVS at rated load.
Consequently, the efficiency of HDCM at rated load is assame as one in TCM, whereas at light load of 0.2 p.u., theefficiency is improved by 1.5% with the HDCM operation.
Therefore, by applying HDCM, the high efficiency isachieved at wide load range, which results in the highweighted efficiency of 98.8% at the switching frequency ofVgs1 (25 V/div)

Vgs1 (25 V/div)

0V
VCds1 (250 V/div)0V
Low turn-off loss

VCds1 (250 V/div)

ZVS of SW1
0V

0V
0V

Vgs2 (25 V/div)

Vgs2 (25 V/div)

0V

Time
(100 ns/div)

Time(100 ns/div) 0V

ZVS of SW2

Low turn-off loss
0V
VCds2 (250 V/div)

VCds2 (250 V/div)(a) ZVS of SW1.
(b) ZVS of SW2.
Fig. 11. Waveforms of gate signal and junction capacitor voltage in HDCM
operation which is the magnified waveforms from Fig. 10(e). By flowing
the TCM current during the zero-current interval of DCM, ZVS isachieved for both switches SW1 and SW2, which results in no turn-on loss.
Furthermore, the turn-off loss can be greatly reduced by connecting asnubber capacitor parallel to the switch in order to delay the increase of thejunction voltage vCds1 and vCds2.

Using the control frequency of 10kHz to control the switchingfrequency of 100kHz is the reason of this small oscillation.

Average Current Command iL_avg* (0.5 p.u./div)

Average Current iL_avg (0.5 p.u./div)

Average Current iL_avg (0.5 p.u./div)
Inductor Current iL (5A/div)
Time (10 ms/div)

0

0A

0

Average Current Command iL_avg* (0.5 p.u./div)

0A
Inductor Current iL (5A/div)

TCM current peak is adjusted according to the zero current period.

Time (10 ms/div)

Conversion Efficiency η [%]Ratio between RMS current andaverage inductor current iL_rms/iL_avg(a) Step-up HDCM current response (0.2 p.u. to 0.4 p.u.).
(b) Step-down HDCM current response (0.4 p.u. to 0.2 p.u.).
Fig. 12. Current response of HDCM. By controlling the positive and negative values of the TCM current, the average current of HDCM is controlled as sameas DCM, which has been research thoroughly. However, the control frequency is required to be the same as the switching frequency in order to eliminate
completely the small oscillation in the average current.
99.0
4.0
HDCM
3.5
98.5
3.0
Maximum Efficiency: 98.9%
Improved by
TCM
1 p.u. of output power is 600W.
1.5%
Reduced2.5
98.0
by 47.2%
1) Efficiency at light load of 0.2 p.u.
2.0
TCM
HDCM: 98.5%
97.5
TCM: 97.0%
1.5
HDCM2) Weighted efficiency (CEC method)
1.0
HDCM: 98.8%
97.0
TCM: 98.6%
Desiredratio
betweenRMS
current
and
0.5
average inductor current iL_rms/iL_avg
96.5
0.0
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1
Output Power Pout [p.u.]
Output Power Pout [p.u.](a) Ratio between RMS current and average current of TCM and HDCM.
(b) Efficiency characteristic of TCM and HDCM.
Fig. 13. Comparison of RMS current and efficiency between TCM and HDCM. In HDCM, by flowing the TCM current during the zero-current interval of
DCM, both ZVS and the current ripple reduction at light load are achieved without any additional components. In particular, the ratio between the RMS
current iL_rms and the average inductor current iL_avg in the HDCM operation is reduced by 47.2% at most compared to the TCM operation. Consequently,
the HDCM efficiency at light load of 0.2 p.u. is improved by 1.5%.

100 kHz. When the switching frequency is further increased,
the high weighted efficiency can still be maintained by optimizing the snubber capacitor, which is used to reduce theturn-off loss, and the TCM current peak, which is designed
to achieve ZVS.
Fig. 14 shows the advantages of the soft switching in
HDCM and TCM over the hard switching in CCM. In the
CCM operation, the hard switching leads to the occurrenceof false triggering due to the high di/dt of the current. Two
factors which increases significantly the di/dt of the current
during the hard switching are as follows; the recovery current of the free-wheeling diode and the discharge current of
the junction capacitor. In order to avoid the occurrence of thefalse triggering during the hard switching in CCM, the main
circuit and the gate driver circuit are required to be customized for the high di/dt of the current, which is undesirable
due to the increase in cost. Consequently, the soft switchingin HDCM and TCM not only reduces the switching loss but
also benefits the design cost of the main circuit and the gate
driver circuit.

Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)
Time (2 ms/div)

14(b)
0V

0V
0A

Soft switching in TCM

Hard switching in CCM

Inductor Current iL (10 A/div)(a) Soft switching in TCM and hard switching in CCM.
Gate Signal of SW1 Vgs1 (10 V/div) Gate Signal of SW2 Vgs2 (10 V/div)
Time (2 ms/div)

0V

V. CONCLUSION
This paper proposed a hybrid current mode between
DCM and TCM, which was entitled Hybrid Discontinuous
Current Mode, i.e. HDCM. In the HDCM operation, by applying the TCM operation into the zero-current interval of
DCM, ZVS was achieved for all the switches without any
modification of the main circuit. Furthermore, the total
charge during the additional TCM interval was neutralized inorder to have no influence on the DCM operation. This enabled the current ripple be reduced at light load in order tomaintain the high efficiency at wide load range. In particular,
compared to the TCM operation, the RMS current at lightload of 0.2 p.u. was reduced by 47.2% with the applicationof HDCM. Consequently, the efficiency of HDCM was improved by 1.5% at most compared to TCM.
In the future works, the application of HDCM in otherconverters, e.g. the single-phase inverter and the three-phase
inverter, will be presented.
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