Novel DC to Single-Phase Isolated AC Converter
using Coupled Inductor with Power Decoupling Capability
Nagisa Takaoka, Keisuke Kusaka, Hayato Higa, Jun-ichi Itoh
Nagaoka University of Technology
1603-1 Kamitomioka-machi
Nagaoka city Niigata, Japan
Tel., Fax: +81 / (258) – 47.9533.
E-Mail: ntakaoka@stn.nagaokaut.ac.jp, kusaka@vos.nagaokaut.ac.jp,
hhiga@stn.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp
URL: http://itohserver01.nagaokaut.ac.jp/itohlab/index.html

Keywords«Converter circuit», «Power converters for EV», «Power conditioning», «Single phase system»

Abstract
This paper proposes a novel active power decoupling converter topology, where additional
components are not necessary in order to minimize the power decoupling circuit. In conventional
EV/PHV battery charger systems with the power decoupling capability require several passive
components and switching devices in order to absorb the power ripple at twice grid frequency. Thus,
these topologies restrict the downsizing. In order to avoid the additional components for the power
decoupling capability, this paper proposes the isolated DC to single-phase AC converter with an
employment of a coupled inductor. This converter consists of a full-bridge inverter, a high frequency
transformer and an indirect matrix converter in order to eliminate DC-link smoothing capacitors.
Moreover, the additional inductor for the power decoupling circuit in the conventional topology can besubstituted with leakage inductor of the coupled inductor in the proposed converter. In the proposed
circuit, the coupled inductor is an isolation between the DC and the AC equipment for safety and theboost inductance. These characteristics result in a high-power-density converter. The validity of the
proposed circuit is experimentally confirmed by a simulation and experiment at prototype. As asimulation result, the proposed method reduces the second-order harmonic of the power ripple by 97.5%
at 3-kW of the output power. As experimental results, the proposed converter reduces the DC current
ripple generated by a single-phase AC load by 46.5%.

I. Introduction
Recently, due to an increasing focus on energy saving, research on PHV/EV has been rapidly
expanded [1-3]. As a main component of a battery charger for these vehicles, a DC to single-phase AC
isolated converter is employed in order to provide a safe connection between DC bus and general singlephase AC equipment. Downsizing and long life-time are required for the DC to single-phase AC isolated
converters. In this converter, DC bus voltage has a power ripple at twice the grid frequency which is
caused by instantaneous power mismatch between the DC side and the single-phase AC grid side. A
large capacitance value of the DC-link capacitor is required in the DC-link to absorb the power ripple
in the conventional method, leading to the use of bulky electrolytic capacitors. However, these bulky
electrolytic capacitors limit life-time and the downsizing of the DC to single-phase AC converters.
In order to solve these problems, many circuit topologies with the power decoupling capability havebeen studied [4-11]. In particular, these circuit topologies employ passive components and several
additional switching devices for power decoupling capability, which is defined as active power
decoupling capability. In such active power decoupling capability, the capacitor voltage is actively
fluctuated, so only small capacitance value is required and replaced with a film capacitor or ceramiccapacitor. Thus, these systems with the active power decoupling capability achieve the long life-time
because the bulky electrolytic capacitors are eliminated. Focusing on [4], the DC to single-phase AC
non-isolated converter is introduced as a simple configuration with the power decoupling circuit
constructed in the third leg of the full bridge inverter. This topology controls the buffer capacitor voltageof the power decoupling circuit using two additional switching devices and an additional inductor. Thisadditional inductor prevents the minimization of the converter.
In order to avoid the additional magnetic component, the DC to single-phase isolated converters withboost-type topologies are also studied. These converters consist of a fly-back-type inverter andadditional switching devices for the buffer circuit. Although the employment of the fly-back-type
topology eliminates the additional inductor, the power capacity of the fly-back-type topology is limited
in term of the converter-efficiency. As a conclusion, the conventional topologies in [4-9] require either
the additional switching devices or the additional inductor for the power decoupling circuit. This leads
to a low efficiency and difficulties in downsizing.
This paper proposes a novel DC to single-phase AC isolated converter using the coupled inductor toachieve the power decoupling capability. A boost converter with coupled inductor is applied in the
proposed circuit because this topology reduces both inductor volume and conduction loss compared toconventional boost topologies [12-13]. Therefore, an interleaved topology of the boost converters withcoupled inductor is suitable to employ as a primary-side converter of the DC to single-phase AC isolated
converter.
The originality of this paper is that the transformer between the DC side and the AC side is integrated
as the coupled inductor in order to eliminate any additional switching devices or any additional inductor
for the power decoupling capability. In particular, the primary-side converter of the proposed system iscomprised of a full bridge inverter, which also operates as two-phase interleaved boost converters toboost the voltage of small capacitors, which absorb the power ripple. Owning to the boost converters,
the buffer capacitor voltage can be pushed up higher than the DC input voltage. Consequently, the buffer
capacitance for the power decoupling circuit is reduced by the fluctuation of the buffer capacitor at highvoltage. Moreover, the additional inductor for the power decoupling circuit in the conventional topology
is substituted with a leakage inductor of the coupled inductor in the proposed converter; meanwhile, theadditional switching devices are unnecessary because the primary-side converters are operated as a
power decoupling circuit during the common-mode of the coupled inductor. On the other hand, thesecondary-side converter can reduce the switching loss because the secondary-side converter is operated
at the twice grid frequency. Thus, the proposed converter achieve the downsizing and the long-lifetimeby the elimination of the additional components.
This paper is organized as follows; first, the proposed converter for the DC to single-phase AC
converter and the principle of the power decoupling method are introduced; second, the operation modeof the proposed converter and the modulation methods with power decoupling capability are explained;
finally, the fundamental operation waveforms of the proposed converter are evaluated in simulations
and experiments.

II. Circuit Topology
A.

Conventional Circuit
Figure 1 depicts a conventional isolated converter for the DC to single-phase AC converter. The
conventional topology consists of an interleaved boost converter, an isolated DC to DC converter andan H-bridge inverter. The interleaved boost converter at the primary-side in the conventional system isrequired in order to push the low input voltage, e.g. the low voltage of the photovoltaic panel, up to thevoltage higher than the peak voltage of the single-phase grid voltage. This two-phase interleaved
topology is employed to reduce the conduction loss and the volume of the boost inductor by dividing
the current by half. In addition, the full-bridge inverter at the primary side of the isolated DC to DC
converter generates a high-frequency square voltage, which can reduce the volume of the transformer.
In this topology, the DC current has a power ripple component caused by the single-phase AC side.
Thus, a bulky electrolytic capacitor Cdc2 is required in the DC-link in order to compensate the power
ripple. However, this capacitor Cdc2 limits the converter’s life-time.

Figure 2 depicts a conventional isolated
igrid
DC-single-phase-AC converter with the
Lboostconventional power decoupling circuit.
Cdc
Lload
This power decoupling circuit is applied at
L
boost
Cdc2
the output of the boost circuit in order tovout vgrid
eliminate the bulky electrolytic capacitor.
idc
In the conventional power decouplingVdc
method, the buffer capacitor voltage isInterleaved
Isolated DC-DC PWM Inverter
actively fluctuated in order to absorb theconverter
converter
power ripple. Therefore, the DC-link
Figure 1. Conventional isolated DC to single-phasecapacitor value can be minimized in
AC converter with interleaved boost topology. The
comparison with that in Fig. 1 because theconventional converter uses a bulky electrolytic
power ripple is compensated by the highcapacitor Cdc to absorb the power ripple caused by a
voltage fluctuation at the small buffer
single-phase load.
capacitance of Cbuf. However, theconventional power decoupling method
Power decoupling circuit
requires additional switching devices andigrid
magnetic components. Consequently, thismethod increases the volume of the cooling
L1
Lload
idc
system and the converter loss.
C
dc1
C
buf
Cdc2
Lboostvgrid
B.
Proposed Circuit
Figure 3 presents the proposed isolated
Vdc
DC to single-phase AC converter. The
Isolated DC-DC
Boost converter
PWM Inverter
proposed converter consists of an indirectconverter
matrix converter and an interleaved boost
Figure 2. Conventional isolated DC to single-phaseconverter with the coupled inductor. The
AC converter with active buffer circuit. The
third winding of the coupled inductor isconventional converter uses a bulky electrolyticelectrically isolated to the primary side inorder to transfer the power to the secondary capacitor Cdc to absorb the power ripple caused by aside. A buffer capacitor Cbuf and a boost single-phase load.
inductor Lboost contributes to the power
ripple compensation. In particular, theS2ap S2bp
charge and the discharge of Cbuf
iload
S1ap S1bp
compensate the power ripple, whereas theN3
Lloadboost-inductor current ibc is controlled by
vdif Cbuf
N1
idc
vs
the
interleaved
boost
converter.
vload
vout
LboostN2
Furthermore, in the proposed converter, thevCbufapplying the three-winding inductor
Vdc vcom
S2an S2bn
S1an S1bn
achieves the DC current control and thegrid current control at the same time. In
particular, the DC current control is
Figure 3.
Proposed isolated DC to single-phase
employed with a common-mode operation AC converter. The primary-side converter togetherof the three-winding inductor, whereas thewith the coupled inductor and the buffer capacitor are
grid current control is implemented with acontrolled in order to convert the DC input power todifferentialmode
operation. the AC power and absorb the power ripple without
Consequently, the system with the power
additional switching devices and inductors.
decoupling capability operates the twocurrent controls without additionalswitching devices and buffer inductors owing to the employment of the three-winding coupled inductor.
The indirect matrix converter in the secondary side is employed to eliminate the DC-link smoothing
capacitors in a conventional Back-to-Back system. The indirect matrix converter performs the rectifier
operation and the polarity reversion of the input voltage to output the grid voltage. Therefore, the DC/AC
conversion of the indirect matrix converter is achieved with the significant loss reduction of theswitching loss because the switching is operated at twice the grid frequency. It is should be noted that

in case of the required bi-directional
operation, a full bridge inverter is appliedinstead the diode rectifier of the indirect
matrix converter.

t

0

t

igrid= 2Igrid sin(ωot)

t

0

＋

t

0

discharge
pdc

pbuf

chargedischarget0

Input power
Pdc [W]0

Buffer powerpbuf [W]pgrid＝ Vload I load 1  cos  2o t 

igrid

＝

pout  2Vload sin o t   2 I load sin o t 

t

t

0

chargeA.

0vgrid= 2Vgrid sin(ωot)

0

ibuf

＝

Principal of power decoupling
Figure 4 depicts the principle of thepower decoupling method in this section.
This power decoupling method is achievedby the compensation of the power ripple in
the DC-link voltage with a small
capacitance Cbuf. If the grid current igrid issinusoidal waveform and the unity power
factor is achieved, an instantaneous output
power pout is expressed by the following:

vgrid

＋

Idc

＋

III. Control Strategy

vCbuf
VCmax
VCmin

Vdc

＝
t

0

t

Output power
pgrid [W]Fig. 4. Relationship among each waveforms.

 Pave  Pave cos  2o t (1)where Vload is the root-mean-square values of the output voltage, Iload is the RMS value of the output
current, Pave is the average output power and ωo is the grid-side angular frequency. The ripple component
shown as the second term on the right side of (1) should be eliminated in order to obtain a constant DC
current in the DC side. Note that other topologies which absorb the power ripple, for example C-L-C
topology, are not effective to reduce the volume of the conventional circuit.
pbuf  Pave cos  2ot (2)

The polarity of the buffer power is defined to be positive when the buffer capacitor is charged. The
buffer capacitor voltage vCbuf which needs to absorb the power ripple is calculated based on the energyof the buffer capacitor. First, the energy of the buffer capacitor WCbuf is expressed by a voltage-current
equation of a capacitor and (2).
dvCbuf t
WCbuf   vCbuf  Cbuf
 d
t0
d 


(3)

  Pave cos  2o  dtt0

where t0 is a start time of operation. By using (3), the buffer capacitor voltage to compensate the power
ripple is presented by the following:vCbuf  VC 0 
2Pavesin2ot   sin2ot0 oCbuf(4)

where Vco is the buffer capacitor voltage at a start time. If t0 is zero, (4) is rewritten as (5).
vCbuf  VC 0 2 Paveo Cbufsin  2o t (5)In the proposed circuit, the DC current flowing to the coupled inductor from DC supply is controlled asa constant DC value. Thus, the buffer capacitor voltage has the power ripple components. The proposed
converter with the power decoupling capability is achieved, whose result is that the power ripple
component at DC side is absorbed.

S1ap

vdif
vcom

Lboost
S1an
Vdc

S1ap

S1bp
Cbuf
VCbuf

vdif
vcom

S1an

S1bn

(b) Differential mode 2

S1ap

S1bp

Lboost
S1an

Cbuf
VCbuf

Vdc

vdif
vcom

LboostS1bn(a) Differential mode 1

S1ap

S1bp

Cbuf
VCbuf

S1bp
vdif

vcom

Lboost
S1an

S1bn

Cbuf
VCbuf

S1bn

Vdc

Vdc
(c) Common mode 1(d) Common mode 2

Fig. 5. Operation modes of primary-side converter with coupled inductor. The primary side of the
proposed system is operated in two modes; the differential mode when the power is delivered to thesecondary-side and the common mode when the power ripple is adsorbed.
B.

Duty calculation
Figure 5 depicts the operation modes of the primary-side converter with the coupled inductor. The
proposed primary-side converter has four operation modes which generate a phase difference between
the switching of each arm as an interleaved converter. The boost inductor Lboost is considered as a leakage
inductor of the coupled inductor. Therefore, if the inductance value of the leakage inductor is insufficientfor the design value, a small inductor is added. In the proposed system, the differential-mode voltage ofthe coupled inductor is controlled to generate the output current, whereas the common-mode voltage iscontrolled to perform the power decoupling capability. In the control of the differential-mode voltage,
the three-winding voltage of the coupled inductor is controlled into a three-level voltage by operation
modes as shown in Fig. 5 (a) and (b). On the other hand, the common-mode voltage vcom is controlled toabsorb the power ripple in the control of the common-mode voltage. In particular, the power ripple
component occurs in the buffer capacitor voltage, because the DC current is regulated into a constantvalue in the proposed power decoupling method. Moreover, the proposed power decoupling capability
consists of a boost topology. The coupled inductor is boosted the fluctuation of the buffer capacitor
voltage in bigger that the DC voltage Vdc. The capacitance of the buffer capacitor is achieved to reduceenough to increase the fluctuated voltage of the buffer capacitor to absorb the power ripple. Thus, theproposed converter with the boost topology of the power decoupling capability is achieved to reduce thecapacitance than a buck-type topology of the one.
In the proposed method, each duties of the primary converter in one carrier period requires to meeta following equation (6).

1  ddif _ a  ddif _ b  dcom _ c  dcom _ d(6)where ddif_a and ddif_b are the duties of the differential mode as shown in Fig. 5(a) and (b), whereas dcom_c
and dcom_d are the duties of the common mode as shown in Fig. 5(c) and (d). If the output voltage iscontrolled into a sinusoidal waveform, the voltage pulses between the first and second winding arecontrolled as PWM pulses. The sum of the differential-mode duties ddif is expressed as follows:d dif  d dif _ a  d dif _ b vout *

2Vdc

2Vload sin(t )
2Vdc

(7)

where vout* is a reference of the output voltage.
Then, the common mode of the proposed method is applied the remaining period after the differential
mode. Therefore, the duty of the common mode is calculated by the following:v *dcom _ c  (1  d dif ) com
2Vdc

(8)

where vcom* is a reference of the common voltage and is output by a PI control of the DC current as
explained in the next section.
Figure 6 depicts a relationship among each reference, the triangle carrier and the switching patternof the primary converter in the proposed system. In the differential voltage mode, the differential voltage
pulses between the first and second winding are controlled as PWM pulses. Then, the voltage the periodof the each duties as shown in Fig. 5(a) and (b) in a carrier period is needed to be equal in order tosuppress a DC biased magnetization of the coupled inductor, especially. The comprising signals A and
B is generated by comprised duties which the sum of dcom_c and ddif and only dcom_c and a triangle carrier
as shown in Fig. 6. Thus, the desired operation ineach duties of the primary converter is actualized as
1
Carrier
described previously. The primary converter of thedcom+ddif
proposed system achieves the power decoupling
dcom
operation and the transfer power to the secondary
0
1side as a same time without the additional passiveA
0
1
B
components and the switching devices.
0
vdif

Control Strategy
Figure 7 shows the control block diagram of the
proposed system with the power decoupling
capability. The current controls and voltage
controls are employed with PI controllers. The DC
current is controlled to a constant DC value toabsorb the power ripple component. This DC
current control is applying as a minor loop of thebuffer capacitor voltage control. This buffer
capacitor control controls only the average of thebuffer capacitor voltage vCbuf_ave in order to avoid
the divergence of vCbuf_ave. The average value of thebuffer capacitor voltage vCbuf_ave is twice of the

0

C.

MODE
S1ap
S1an
S1bp
S1bn

2

vcom*
Fbuf (s)

PI

1
2Vdc

×

+
-

1
Carrier

igrid*

D2
0
1
1
0

C2
0
1
0
1

Normalization of voltage reference

Filteridc
DC current control

vCbuf
Average voltage
control for Cbuf

C1
1
0
1
0

vout*
PI

1
2Vdc

igrid
Current control of output filter inductor+

Pulse patternconversion

PI

D1
1
0
0
1

Fig. 6.
Current and voltage operating
waveforms of proposed converter with power
decoupling capability. The buffer capacitor
voltage absorbing the ripple component is
controlled by the zero-voltage period of the
primary-side transformer voltage.

idc*
vCbuf_ave*

C2
0
1
0
1

polarity
conversion

S1ap
S1an
S1bp
S1bn

vdif*
S2ap
S2an
S2bp
S2bn

Fig. 7. Proposed control block diagram. The DC current is controlled to absorb power ripple. The
buffer capacitor voltage control is employed in order to avoid the divergence of the average voltage
due to discretization.

TABLE ISYSTEM PARAMETERS.
DC bus voltage

200 Vdc

Load voltage

200 Vrms

Rated power

3 kW

Load frequency

50 Hz

Boost L (Lboost)
Buffer C (Cbuf)

1.1 mH
150 mF

Filter L (Lload)

2.0 mH

Load current

15 Arms

Turn ratio of coupledinductor N1:N2:N3

1:1:2

Carrier frequency offull bridge inverter
Natural angular frequency
of DC current control
Damping factor of
DC current control

Natural angular frequencyof buffer C voltage control
Natural angular frequencyof load current control

50 kHz6000 rad/s
0.7

30 rad/s
4000 rad/s

Damping factor of buffer
C voltage control

common voltage vcom. In the proposed converter, theoutput current is controlled by the full bridge inverterat the primary side. Then, the diode bridge rectifiers
the differential-mode voltage, whereas the full bridge
inverter at the secondary side operates a unfolder andchanges the polarity of the voltage pulse in
responding to the polarity of the grid voltage. Thus,
the secondary-side converter achieves the reductionof the switching loss.

1.0

DC input voltage Vdc [V]
200

40
20
0
-20
500

DC input current idc [A]Average current

Output voltage vout [V]With LPF

0
-500
450

Buffer capacitor voltage vbuf [V]400

IV. Simulation and Experimental
Results
A. Simulation results in steady state
The performances of the proposed circuit areverified by simulation and experiment.
Table I shows the simulation condition for the
proposed isolated DC-single-phase-AC-converterwith the power decoupling capability. It should be
noted that in the confirmation operation without thepower decoupling capability, the simulation andexperimental results of the power ripple component
is acquired under the conditions that the gain of theDC current controller is reduced and the gain of theaverage voltage controller of the buffer capacitor Cbuf
is increased.
Figure 8 and Figure 9 show the input and theoutput waveforms with/without the proposed power
decoupling method, respectively. A input current isfiltered with a LPF with a cut-off frequency of 1 kHz
in order to remove a switching ripple. Fig. 8 shows
the result without the power decoupling method.
Thus, the input current has a power ripple component
at 100 Hz because the power ripple component is notcompensated by buffer capacitor in this operation
mode. Then, the ripple component is 127.1% withreference to an average current in Fig. 8. In contrast,
as shown Fig. 9, the proposed power decoupling

350
10

Buffer current ibuf (LPF)[A]

0
-10
0.24

0.28 [s]0.26

Fig. 8. Waveforms without proposed power
decoupling method in steady state.
DC input voltage Vdc [V]200

DC input current idc [A]Average current

Output voltage vout [V]With LPF

16
14
12
500
0
-500
450

Buffer capacitor voltage vbuf [V]

400
350

Buffer current ibuf (LPF) [A]
5
0
-5
0.16

0.18

0.20 [s]Fig. 9. Waveforms with power decoupling
method in steady state. The proposed power
decoupling method reduces the power ripple
component at 100 Hz on the input current by
97.5%.

Buffer capacitor voltage vCbuf 100 V/div
Buffer capacitor voltage vCbuf 50V/div.

DC input voltage Vdc 100 V/div
DC input voltage Vdc 50V/div.

Common voltage vcom 250 V/div

Output current iout 2A/div.
0
0

0

0

0

Output voltage vout 100V/div.

4 ms/div.

Differential voltage vdif 250 V/div(a) Fundamental waveforms at 150W

4 μs/div(b) Expansion of each voltage

Fig. 10. Waveforms with power decoupling method in steady state. The zero voltage period of vdif isagreed with the Vdc period of vcom.

Buffer capacitor voltage vCbuf 50V/div.

Buffer capacitor voltage vCbuf 50V/div.

DC input voltage Vdc 50V/div.
DC input current idc 5A/div.

DC input voltage Vdc 50V/div.
DC input current idc 5A/div.

0

0

0

0

Output voltage vout 100V/div.

4 ms/div.

Output voltage vout 100V/div.

4 ms/div.

(a) Without proposed power decoupling method (b) With proposed power decoupling method
Fig. 11. Experimental waveforms with/without power decoupling method in steady state.
method is controlled to actively fluctuate the buffer capacitor voltage vcbuf. Consequently, the secondaryorder harmonics in the DC input current is suppressed greatly. As a result, the ripple of the DC input
current is suppressed by 97.5%.
B. Experimental results
Figure 10 (a) depicts the experimental waveform of the input and output voltage and Figure 10 (b)depicts the extended waveform of the coupled inductor in the proposed converter with open-loop controlof the load current control. It should be noted that these experimental results are obtained with a natural
angle frequency of the input current control of 4000 rad/s, a buffer capacitor of 200 μF, vout of 100 Vrms
and an R-L load as the single-phase AC load. The boost operation, which pushed the voltage of thebuffer capacitor higher than that of the DC voltage Vdc, is confirmed by Fig. 10 (a). On the other hand,
as shown Fig.10 (b), the differential-mode voltage vdif is a three-level voltage waveform, which is
combined of + vCbuf, - vCbuf, and a zero voltage period. This zero voltage periods of vdif in the power ripple
operation is decided by the capacitance of Cbuf and the output power. Moreover, the period of the zerolevel in the differential voltage is utilized with the top and bottom level of the common mode voltage.
Thus, the power decoupling operation and the output current control are achieved at the same time.
Through these results, the operation of the primary-side converter with the power decoupling operationin the proposed system is confirmed without additional switching devices and magnetic components.
Figure 11 shows the waveforms of the proposed converter with/without the proposed power
decoupling method, respectively with Vdc of 150 V and vout of 100 Vrms. As shown in Fig.11 (a), the DC
input current has a power ripple component at second-order harmonics of the grid frequency. The bufferHarmonic component[%](100%: DC current component)

70

Without the power decoupling method

60
50Reduced by 46.5%

40
30With the power decoupling method

20
10
0
1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20
Harmonic number

Fig. 12. Harmonic analysis of DC input current.
capacitor voltage fluctuation is small because the power ripple component is not compensated by buffer
capacitor in this operation mode, i.e. the passive power decoupling capability. Therefore, the capacitanceof the buffer capacitor Cbuf is smaller than the necessary capacitance to absorb the power ripple. In
contrast, from Fig. 11 (b), the proposed power decoupling method is controlled to actively fluctuate thebuffer capacitor voltage vCbuf. Consequently, the second-order harmonics in the DC input current issuppressed greatly.
Figure 12 shows the harmonic analysis of the input current. Note that the fundamental harmonic isbased on the grid frequency component. The harmonic components of the input current up to 20th-orderof the grid frequency are considered in this evaluation. It is confirmed from the analysis result, the ripple
component (100 Hz) without the power decoupling method is 65.6%, which is normalized by the DC
component of idc. In contrast, the proposed method reduces the second-order harmonics of the input
current as shown in Fig. 11 (b) by 46.5%. Consequently, the experimental results verified thefundamental operation of the proposed power decoupling method. The input current ripple will be
reduced more by improving the input current control performance and clarifying the optimized design
to absorb the power ripple component.

V. Conclusion
This paper proposed the DC to single-phase AC isolated converter with the boost-type topology ofthe power decoupling capability by the three-winding coupled inductor. The proposed system with theindirect matrix converter is expected to achieve down-sizing compared to the conventional rectifierinverter system, because in the proposed system a bulky electrolytic capacitor for DC-link is not required.
Moreover, the additional switching devices and magnetic component for the power decoupling operation
do not required in the proposed method. In particular, leakage inductors of the coupled inductor are
employed for the power decoupling circuit and the boost operation of the full bridge inverter. As asimulation result, the power ripple component as a secondary-order harmonics is reduced by 97.5% incomparison without the proposed power decoupling method. In addition, the experimental result verifies
the proposed power decoupling method and the secondary-order harmonics caused by the power ripple
reduced by 46.5%.

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