Highly-Reliable Fly-back-based PV Micro-inverter
Applying Power Decoupling Capability without
Additional Components
Hiroki Watanabe, Nagaoka University of technology, Japan, hwatanabe@stn.nagaopkaut.ac.jp
Jun-ichi Itoh, Nagaoka University of technology, Japan, Itoh@vos.nagaopkaut.ac.jp

Abstract
This paper discusses a verification of an electrolytic
capacitor-less PV micro-inverter aiming for highreliability and volume reduction. The inverters,
which are connected to a single-phase AC grid,
have a double-line-frequency power ripple. Thus, abulky electrolytic capacitor is generally used in DC
side. The proposed converter consists of the flyback converter, the voltage source inverter (VSI),
and small capacitor in the DC link part in order toachieve the power decoupling without the any
additional component. In addition, the sensor lessmagnetizing current control is proposed in order toensure the compatibility between the surge voltage
and cost reduction. Furthermore, it is demonstrated
that the discontinuous current mode (DCM) of the
proposed fly-back converter has the great power
decoupling effect in comparison with the continuous
current mode (CCM). From the experimental result,the second-order harmonics in the DCM operationis reduced by 97% in comparison with that in the
CCM operation.

1. Introduction
Recently, a photovoltaic (PV) system is actively
researched as a sustainable power solution due tothe attractive characteristics such as; flexibility,
high-system efficiency, and low manufacturing cost.
Thus the utilization of the micro-inverters
promisingly become a trend for the future PV
system instead of using large capacity inverters. In
particular, the micro-inverter with the high reliability
is required because a large number of converter
units are adopted to the PV generation system.
However, in the conventional PV system, theelectrolytic capacitors are usually employed owing
to the requirement of the large capacitance. These

capacitor limits the life-time of the converter, which
results in low reliability.
On the other hands, in the converter topologies forthe micro-inverter, the fly-back converter has been
studying for following advantages; (i) Simple
configuration and low manufacture cost (ii) galvanic
isolation (iii) high voltage boost-up (IV) small
volume. In this case, the configuration with the flyback converter and the unfolding bridge is thecenter of attention because the unfolding bridge isswitched at the line-frequency and the switchingloss can be reduced drastically in comparison withthat of two-stage isolated DC/AC converter.
However, the large electrolytic capacitor is requiredin order to compensate the double-line frequency
power ripple owing to single-phase AC grid. As aresult, the life-time of the converter is limited due tothe large buffer capacitor, which results in low
reliability.
In order to solve this problem, the active power
decoupling topologies for electrolytic capacitor-lessconverters have been researched actively [1]-[8].
The active power decoupling can reduce thecapacitance for the double-line frequency power
ripple compensation, which enables to use of film
or ceramic capacitors instead of the electrolytic
capacitor. However, the additional components
such as the switching device and the passive
components are required. Although the low cost is
the one of the advantages of the fly-back microinverter, these components increase complexity of
the circuit configuration and the cost.
This paper presents a simple configuration and thecontrol method of the active power decoupling
without any additional components. The proposed
converter consists of the fly-back converter, the VSI
and small buffer capacitor, which ensures the highreliability. In addition, the sensor less magnetizing
current control is proposed in order to achieve thepower decoupling capability. Owing to the proposed

H.F.
control, the PV input current and the input powertransformer
Small decoupling capacitor (50 mF) with 300W
S2
igrid
become constant. Furthermore, current sensor less
N1:N2 1 : N D1
IPVcontrol contributes the cost reduction.
D1
vgrid
This paper is organized as follows; first, theCbufconfiguration of the conventional micro-inverter with
Vdc
Filter
S1
Cbuf
D2
fly-back converter is explained. Next, the proposed VPVS1converter and a control block diagram are
S3
Filterdescribed. After that, the operation mode of the flyFly-back converter
VSI
iS1
back converter is considered focus on the power
Fig. 4. Proposed
converter
withdecoupling
fly-back converter.
Large capacitor
for power
decoupling. Finally, the fundamental operation ofsmall configurationbuffer capacitor
Cbuf compensates
the proposed converter is demonstrated by the Fig. The
1. Typicallyof fly-backmicro-inverter.thedouble-line frequency power ripple. Thus, thesimulation and experiment. From the experimental
component for power decoupling is not
result, the second-order harmonics is reduced by iS1 additional[A]
required.
Un filtered
97% by the DCM operation. In addition, it isconfirmed that the sinusoidal inverter output current.
Input power p FilteredBuffer power p
in

2. Conventional fly-back microinverter topology

igrid [A]

DC

buf

DC

＋

Figure 1 shows the typical configuration for the
micro-inverter topology with a fly-back converter,DC
AC
and Figure 2 shows the fundamental operation
waveforms of Fig.1. The primary side switch S1 isswitched at the high frequency in order to reduce
Fig. 5. Decoupling approach of the proposed
converter.
the volume of the transformer. In this control, theduty command is adjusted to obtain absorb v [V]
gs
vCbuf
vgrid
Vin
waveform with the grid frequency. After that, theS2secondary switch S2 and S3 is switched at the utility
single-phase AC grid half cycle. On AC side, the
S3 t
0
0
t
0
t
small CL filter is connected in order to eliminate theswitching frequency component. In this case, the Fig. 2. Operation waveforms of typical fly-back microibuf
igrid PWM ismodulation withswitching loss of S2 and S3 can be reduced inverterIinin DCM. Sinusoidal
drastically because the switching frequency of the applied on the primary side MOSFET, and the unfolder
at the line-frequency.
0
0
t
t
t
S2 and S3 is the same as the utility single-phase bridge S0 2 and S3 is switched
grid frequency.
Auxiliary circuit
One of the strong advantages in this configuration
pbuf
pgrid
pin
is a small number of the main circuit components.
chargeHowever, the bulky electrolytic capacitor Cbuf isdischargeCbuf
required in the primary side of the transformer
U.F
because the double-line frequency power ripple
Capd
0
0
t
t
0
t
comes from the single-phase AC grid.
Figure 3 shows the micro-inverter topology with the
Buffer power
Output power
Input power
conventional active power decoupling circuit [9]. In
pbuf [W]pout [W]pin [W]
U.F
Cbuf : smallorder to reduce the capacitance of Cbuf, the small
Fig. 6. Principle of the power ripple compensation. In
buffer capacitor Capd and the auxiliary circuit areorder to obtain constant input power Pin, the buffer
added in the transformer secondary side. In thiscapacitor voltage Cbuf is fluctuated at the twice gridcase, the Capd is charged and discharged in Fig.frequency.
3. Conventional fly-back micro-inverter with active
synchronized with the double-line frequency power power decoupling. It can reduce the capacitance ofripple. As a result, the DC side capacitor Cbuf Cbuf. However, auxiliary circuit is required, andbecomes small. However, these additional expensive.
components complex the circuit configuration. In
addition, the manufacturing cost will be increased＝

by the additional component for active power
decoupling.

Small decoupling capacitor (50 mF) with 300W
N1:N2
D1

3. Proposed converter

Vdc

Figure 4 shows the proposed converter which
consists of the fly-back converter, the VSI andsmall buffer capacitor Cbuf. The Fly-back converter
isolates between the PV and the single-phase grid,and boosts the input voltage. After that, PV panel isconnected to the single-phase grid by VSI. In the
proposed converter, the double-line frequency
power ripple is compensated by the Cbuf, and the
additional components such as the switching
device and the passive components are notrequired in order to achieve the power decoupling.
Thus, the proposed converter can reduce the cost
and the volume of the converter in comparison withthat of the conventional fly-back micro-inverter withthe active power decoupling circuit as shown in Fig.
3.
Figure 5 shows the power decoupling strategy ofthe proposed converter. The unmatched power
between the input and output power iscompensated by the DC link capacitor. Figure 6
shows the principle of the power decoupling
between the DC and single-phase AC sides. Whenboth the output voltage and current waveforms are
sinusoidal, the instantaneous output power pout isexpressed as

Vacp I acp(1  cos 2t )2FilterFly-back converter

VSI

Fig. 4. Proposed converter with fly-back converter.
The small buffer capacitor Cbuf compensates the
double-line frequency power ripple. Thus, theadditional component for power decoupling is not
required.
Input power pin

Buffer power pbuf

DC

DC

DC

AC

Fig. 5. Power decoupling strategy of the proposed
converter.
vCbuf

Vin

0

t

vgrid

t

0

ibuf

Iin

igrid

(2)

where the polarity of the pbuf, is defined as positive
when the buffer capacitor Cbuf discharges. Note thatthe active power of Cbuf should be zero. Owing to
the power decoupling capability, the input power ismatched to the output power. Thus, the relationshipbetween the input and output power is expressed as

pbuf

pin

0

t

0

pgrid

chargedischarge

t

0

t

t

0

0

＝

where Vacp is the peak voltage, Iacp is the peakcurrent, and  is the angular frequency of the output
voltage. From (1), the power ripple that contains
double-line frequency of the power grid, appears at
DC link.
In order to absorb the power ripple, theinstantaneous power pbuf should be controlled by

t

t

0

(1)
0

1
pbuf  Vacp I acp cos 2t
2

Cbuf

＋

pout 

S1t

Buffer power
Output power
Input powerpbuf [W]pout [W]pin [W]
Fig. 6. Principle of the power ripple compensation.
In order to obtain constant input power Pin, the buffer
capacitor voltage Cbuf is fluctuated at the twice gridfrequency.

1pin  Vacp I acp  VPV I PV
2

(3)

Vgs_s1

4. Operation mode of the fly-backconverter focus on power
decoupling capability

Id_s1

ID

Figure 7 shows the primary current waveforms of
the CCM and the DCM. When the MOSFET S1 isturned on, the magnetizing inductor Im is charged.
When the S1 is turned off, the inductor energy istransferred to the secondary side through the diodeof D1.
Where, the average primary current both CCM and
DCM is expressed as

Tsw(a) CCM operation.
Vgs_s1

Vdc 
I dc
Vin



(4)

Id_s1

I ave _ DCM 

I peak

Don
2



(5)

ID



(6)

Vin
DonTsw 
Lmwhere, Iave_ccm is the average primary current in
CCM, Vdc is the DC link voltage, Idc is the secondary
input current, Vin is the PV input voltage, Lm is themagnetizing inductance, Don is the on-duty of S1,
Tsw is the switching period.
Figure 8 shows the waveforms of the DC link
voltage Vdc and DC link current Idc. From (4), in the
CCM operation, the primary average current Iave_ccm
is decided from the Vdc and Idc. However, the Vdc
and Idc are fluctuated owing to the double-line
frequency power ripple. Therefore, the cut-off
frequency of the current regulator, which is enoughhigher than double-line frequency, should bedesigned. As a result, the Iave_ccm is fluctuated, andit means the power decoupling is not achieved
perfectly.
On the other hand, in the DCM operation, theprimary average current is decided from theprimary side parameter only from (6). Thus, theoutput side condition is not affected to the PV input
side. As a result, the power decoupling operation isachieved consistently in the DCM operation. For
these reason, the proposed circuit can apply thepower decoupling operation in DCM with very
simple operation.

DonTsw

DoffTsw

I ave _ CCM 

I peak 

Iave_CCM (Iin)

Ipeak
Iave_DCM (Iin)

DonTsw

DoffTsw

Tsw(b) DCM operation.
Fig.7. Primary current waveform.
Idc

N1:N2
D1
Vin

S1

Pgrid
Vdc

VSI

Vdc

Idc*VdcIdc (filtered)

Idc*Vdc is matched tothe output power Pgrid

Fig. 8. Waveform of DC link voltage and current. In
CCM operation, input power is affected to output
power ripple because primary current is decided by
DC link voltage and current.

+

Im*

5. Control block diagram－
Im_estDetection

Figure 9 shows the control block diagrams of the
proposed converter. In the fly-back control, themagnetizing current control is applied for power
decoupling. When the fly-back converter isoperated in CCM, the double-line frequency power
ripple should be compensated in order to achieve
the power decoupling. Note that the angularfrequency of the power ripple is expressed as power _ ripple  4f grid

vdc* +(9)

Note that, when the fly-back converter is operated
in DCM, the acr can design low because the input
power in the DCM is decided by only duty ratio of
the switch S1. Thus, in the DCM, the acr isdesigned by the maximum power point tracking(MPPT), and it has the low response in comparison
with the power_ripple.
On the other hand, the proposed current control
needs the parameter of the magnetizing current Im.
However, the current sampling is complicatedbecause the Im has non-linear characteristics due tothe flux saturation. In addition, when the wire
inductance becomes high, it leads to the large
surge voltage in the switch S1. In order to solve thisproblem, the current sensor less control isdeveloped.
Firstly, the ratio between the input and the output
voltage of the fly-back converter is expressed as
Vdc N 2 dVin
N1 1  dPI

2 sin q(10)where, Vdc is the DC link average voltage, Vin is the
input voltage, N1 and N2 is the number of turns of
the transformer, d is the on-duty of the switch S1.
According to (10), d is expressed as (11)

PI

+

+
+

+
vdc

BEF

(8)
PLL

q

1

Sup
Sun

igrid
-1

-1

where, fgrid is the grid frequency. In order to obtainthe constant input power in CCM, the current
command of the magnetizing current control Im* isgiven by the constant value. In order to suppress
the power ripple, the angular frequency of thecurrent controller has to satisfy as (9)


S1

-

vgrid

vgrid

 power _ ripple  acr 

Carrier0(a) Fly-back converter control

vdc

Magnetizing
Current
Estimation
Eq. 13

VPV
IPV
Vdc

1 Duty
+
Vdc

PI

+
-

Swp
Swn

(b) VSI control
Fig. 9. Control block diagram.

d1
Vin N 2
1 
Vdc N1(11)

On the other hand, the magnetizing current Im isexpressed as
Im 

1 N2
I dc 
1  d N1(12)where, Idc is the DC link current.
Finally, the estimation value of the magnetizing
current Im_est is expressed to (13) from (11) and (12)VPV
1
PPV N 2 Pin
N 2 Vdc
I m _ est (  I in ) 
N1 VPV Vdc
N1 VdcVdc(13)

In expression (9), the Idc is calculated from the input
power PPV and the DC link average voltage.
In the VSI control, the inverter output current
control (ACR) and the DC link voltage control(AVR) are implemented in order to connect to thesingle-phase AC grid, and the DC link voltage
command Vdc* in AVR is set more than the peak
grid voltage Vacp. Note that, the DC link voltage has
the double-line grid frequency component owing to
the proposed decoupling control. As a result, thetotal harmonic distortion (THD) of the inverter

output current is decay. In order to solve thisproblem, a band eliminate filter is applied to the DC
link voltage detection. The AVR controls the Vdc byreferring to the only average value of the DC link
voltage.
Finally, the phase looked loop (PLL) is applied toensure that the phase angle of the inverter output
current is identical to the single-phase AC grid.

Table.1 Simulation condition.
Symbol
VPV
Pout
fsw
Cbuf
Lm
vac
fac

Quantity
value
Input voltage
50 V
300 W
Output power
Switching frequency
80 kHzDC link capacitor
50 mF
Magnetizing inductor 5 mH(200 mH)
Grid voltage
200 Vrms
Grid frequency
50 Hz

6. Simulation results
Input voltage VPV

Table 1 shows the simulation parameters. Figure
10 shows the simulation results with the CCM
operation. In the simulation, the operation of MPPT
is not considered. The rated power is set to 300 W,
and the buffer capacitor is 50 mF. In addition, thegrid frequency is 50 Hz, and the angular frequencyof the power ripple power_ripple is 625 rad/s.
According to Figure 10 (a) and (b), when the rate
between the angular frequency of the ACR acr andthe power_ripple is low, the PV input current isfluctuated at the double-line grid frequency. On theother hand, when the acr is increased to 4000
rad/s, the fluctuation of the PV input current
becomes small. From these results, when the flyback converter is operated in CCM, the acr should
be set more than the power_ripple in order to achieve
the power decoupling capability. In addition, theinverter output current becomes sinusoidal
waveform, and the buffer capacitor voltage Vdc isfluctuated at the double-line grid frequency owing to
the power decoupling. From these results, thevalidity of the fundamental operation of the
proposed converter is confirmed.
Figure 11 shows the simulation results with the
DCM operation. According to the Fig. 10 (a), whenthe acr is set to 1000 rad/s, the PV input current
has the double-line grid frequency component of
16.9%. However, According to the Fig. 11, the PV
input current fluctuation is less than 1%. Following
this result, it was confirmed that the power
decoupling effect in DCM is larger than the CCM
operation.
Figure 12 shows the comparison with the input
current second-order harmonics with both the CCM
and the DCM. In this case, the acr is set to4000rad/s. According to Fig.12, the second-orderharmonics in DCM is less than the CCM operation.
This is because the output side condition is notaffected to the PV input power as shown in chapter
4.

[V]
Input current IPV[A]

Fluctuation
Capacitor voltage vdc[V]
Grid voltage vgrid[V]
Inverter output current igrid[A]
10msec/div(a) acr : 1000rad/s
Input voltage VPV[V]
Input current IPV[A]Constant
Capacitor
Capacitorvoltage
voltagevvcbuf
dc[V]
Grid voltage vgrid[V]
Inverter output current igrid[A]
10msec/div(b) acr : 4000rad/s
Fig.10 Simulation results in CCM operation.
When the angular frequency acr increases, theinput current becomes constant.

In order to demonstrate the validity of the proposed
decoupling method, a 300 W class prototype circuit
is tested. Table 2 shows the experimental
parameters, Figure 15 shows the experimental
results with the CCM and DCM operation. In thisexperiment, the sensor loss current control is notapplied because the purpose of this experiment is
the comparison of the decoupling effect between
the CCM and DCM operation. In addition, in order toclarify the difference between them, the fly-backconverter is operated in the open loop control.
According to Fig. 14 (a), when the fly-backconverter is operated in CCM, the input current isfluctuated at the double-line grid frequency. This isbecause the input current is affected from the output
power ripple. It means the power decoupling is notachieved.
According to Fig. 14 (b), when the fly-backconverter is operated in DCM, the input current
fluctuation is reduced drastically in comparison with
the CCM operation. From these results, the DCM
operation of the fly-back converter has a power
decoupling effect more than the CCM operation.
Note that, this power decoupling operation is limited
to the constant duty condition.
Figure 15 shows the comparison with the secondorder harmonics on the input current. In the DCM
operation, the second-order harmonics is reducedby 97% in comparison with that of the CCM
operation. From these results, the validity of the
proposed power decoupling is confirmed by thisexperiment.

Conclusion
This paper presents a simple configuration and thecontrol method of the active power decoupling
without any additional components for fly-backmicro-inverter. In order to achieve the power
decoupling, the magnetizing current control is[V]
Input current IPV[A]Constant
Capacitor
Capacitorvoltage
voltagevvcbuf
dc[V]
Grid voltage vgrid[V]
Inverter output current igrid[A]
10msec/div

Fig.11 Simulation result in DCM operation whenacr is 1000rad/s. in the DCM operation, highspeed response of acr is not required.
2.5

Input current secondary-orderharmonics [%]7. Experimental results

Input voltage VPV

100% = DC
Fundamental frequency : 50 Hz

2.0
CCM(acr = 4000rad/s)

1.5
1.0

DCM
(acr = 4000rad/s)

0.5
0.0

0

50

100

150
200
250
Output power [W]

300

350

Fig.12 Comparison with the Input current secondorder harmonics both the CCM and DCM.
100

Input current secondary-orderharmonics [%]Figure 13 shows the characteristics of the power
decoupling effect when the angular frequency of
ACR acr was changed. According to Fig.14, in the
CCM operation, the second-order harmonics in the
input current increase when the acr is low. On theother hand, in the DCM operation, second-orderharmonics is small at the all condition in comparison
with the CCM operation. From these results, theDCM operation is validity in order to achieve thepower decoupling.

10

CCM

1
0.1
0.01

Reduced by 94%
DCM

0.001
500 1000 1500 2000 2500 3000 3500 4000 4500
Angular frequency of ACR acr [rad/s]

Fig.13 Characteristics of the power decoupling
effect when the angular frequency of ACR waschanged.

proposed. In addition, the operation mode of the flyback converter for the power decoupling capability
is explained. As a result, it was confirmed that theDCM operation has great effect for the power
decoupling in comparison with that of the CCM
operation. From the experimental result, thesecond-order harmonics is reduced by 97% incomparison with the CCM operation.
In the future work, some characteristics such as theefficiency and the converter loss will bedemonstrated.

Table 2 Experimental parameter

Symbol
Pout
fsw
Cbuf
Lm
Load
fac

Quantityvalue
50 W
Output power
Switching frequency
80 kHzDC link capacitor
40 mF
Magnetizing inductor 13 mH(DCM)
200 mH(CCM)
R-L load
Output frequency
50 Hz

Input voltage Vin [25V/div]

Acknowledgment

Input current Iin [A/div]

This study was supported by New Energy andIndustrial Technology Development Organization(NEDO) of Japan.

Fluctuation

Inverter output voltage Vout 250 [V/div]

Reference
4msec/div

Inverter output current Iout 2 [A/div](a) CCM operation.
Input voltage Vin [25V/div]
Constant

Input current Iin [A/div]

Inverter output voltage Vout 250 [V/div]

4msec/div

Inverter output current Iout 2 [A/div](b) DCM operation.
Fig. 14 Experimental results in order to evaluate
power decoupling effect between CCM and DCM.
100
Input current Iin [%][1] H. Renaudineau, S. Kouro, K. Schaible and M.
Zehelein: “Flyback-based Sub Module PV Microinverter”,
EPE`16 ECCE Europe, (2016)[2] R-K. Surapaneni, A-K. Rathore: “A novel singlephase isolated PWM half-bridge microinverter for solar
photovoltaic modules”, ECCE US pp. 4550-4556 (2015)[3] E. Fonkwe, J. Kirtley, J. Elizondo: “Elyback microinverter with reactive power support capability”, IEEEth
17 Workshop on Control and Modeling for PowerElectronics (COMPEL), pp.1-8,(2016)[4] F. Ji, L. Mu, G. Zhu: “A novel Multi-function
photovoltaic Micro-inverter and its control strategy”,
thIEEE 8 International Power Electronics and Motion
Control Conference (IPEMC-ECCE Asia), pp.13021305,(2016)[5] S-M. Tayebi, C. Jourdan, I. Batarseh: “Dynamic Dead
Time Optimization and Phase-Skipping Control
Techniques for Three-Phase Micro-Inverter Applications”,
IEEE transactions on Industrial Electronics, pp.110,(2016)[6] Yoshiya Ohnuma, Jun-ichi. Itoh: “A Single-Phase
Current-Source PV Inverter With Power Decoupling
Capability Using an Active Buffer”,IEEJ transactions ,Vol. 51, No. 1, pp. 531-538 (2015)[7] X. Liu, M. Agamy, D. Dong,M. Harfman-Todorovic, LGarces: “A low-cost solar micro-inverter with softswitching capability utilizing circulating current”, IEEE
Applied Power Electronics Conference and Exposition(APEC), pp.3403-3408,(2016)[8] Y-M Chen, C-Y Liao: “A PV Micro-inverter with PV
Current Decoupling Strategy”, IEEE transactions on
Power Electronics, pp.1-14, (2016)

CCM

DCM
58%

1.8%

80
60
40
20
0
DC

Second-order
Harmonics(100 Hz)

Fig. 15 Comparison with second-order harmonics
on input current between the CCM and DCM.