Isolated Three-Phase AC to DC Converter with
Matrix Converter Applying Compensation for
Voltage Error by Voltage-based Commutation
Satoshi Nakamura
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japans173164@stn.nagaokaut.ac.jp

Hiroki Watanabe
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japanhwatanabe@vos.nagaokaut.ac.jp

Kashin Kiri
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japans101622@stn.nagaokaut.ac.jp

Jun-ichi Itoh
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
itoh@vos.nagaokaut.ac.jp

Abstract— This paper proposes a compensation method for
a voltage error by a voltage-based commutation of an isolated
AC-DC converter using the matrix converter with a
discontinuous current mode (DCM) and a continuous current
mode (CCM) for rapid battery charger applications. The
isolated AC-DC converter with the matrix converter is expected
to be more compact because a medium-frequency AC inductor
which is connected to the medium-frequency transformer isused instead of a boost inductor. However, the small mediumfrequency inductor causes the input current distortion. In
addition, the transmission power error occurs due to the voltage
error by the commutation. In this paper, the voltage error
caused by the voltage-based commutation is analyzed based on
the relationship between the current direction and the threephase voltage in the initial step of the voltage-based
commutation. The two voltage compensations method by the
two-level compensation and the linear approximation are
combinated. As a result, the input current THD was reduced by
70 % at the boost ratio 1.0.
Keywords— Three phase AC-DC converter, Matrix converter,
Rapid battery charger

I. INTRODUCTION
Recently, electric vehicles (EV) and plug-in hybrid
vehicles (PHV) may become a trend of automotive
technologies as solutions for the energy and environmental
problems. The rapid battery chargers are key component toenhance the convenience for users. The rapid battery chargers
have important requirement of the reduction for the weight
and the size for easy installation and dissemination. Typically,
the main power circuit for the rapid battery charger is amedium-frequency isolated AC-DC converter with a PWM
rectifier. This conversion method requires boost inductors in
the grid AC stage, but it is miniaturization by the highfrequency operation [1]. Whereas, a bidirectional AC-DC
converter which has the matrix converter and the mediumfrequency transformer has been studied actively[2]-[4]. This
circuit directly converts from the three-phase AC to themedium-frequency single-phase AC by the matrix converterin grid side. Thus, the efficiency is improved by reduction of
a number of conversion stage without a DC-link part. In
addition, it has advantage for the size reduction because a

XXX-X-XXXX-XXXX-X/XX/$XX.00 ©20XX IEEE

Shunsuke Takuma
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
takuma_s@stn.nagaokaut.ac.jp

large grid-tied inductor and a DC-link capacitor are notnecessary.
Some control methods for the matrix converter in theisolated AC-DC converter have been proposed [2]-[9]. In Ref.
[2], the switching pattern based on space vector modulation(SVM) is investigated to neglect the effect of the current ripple
within one switching cycle. An inductor current is assumed tobe constant during one switching cycle. However the large
current ripple is occur in the input current because themedium-frequency AC inductor is small. The ripple current
gives an error to the current command value because theconventional SVM is based on constant output current during
the switching periods. As a result, the input current distortion
occurs due to the large ripple current in one switching cycle.
Ref. [4] has proposed a switching method that is not affectedby current ripple caused by leakage inductance in theswitching cycle. However, the current ripple is not canceled
strictly because the actual output period of the voltage vector
differs due to the voltage error due to the commutation. In thisresult, the distortion occur in the input current. In Ref. [5], thevoltage error is eliminated by adjusting the duty ratio
according to the dead-time period when the voltage error isalways constant. However the nonlinearly of the voltage error
causes the overcompensation when the commutation
sequence starts near zero current. The three-levelcompensation, which includes the zero-voltage period,
improves the overcompensation problems. However the threelevel compensation has a trade-off relationship between thevoltage error due to the commutation and the conversion
efficiency.
In this paper, the compensation method of the voltage error
is proposed utilizing the voltage-based commutation of thematrix converter. The new contribution of this paper is that thevoltage error caused by the voltage-based commutation is
reduced by the combination between the two-levelcompensation and the linear approximation compensation
focusing on phase difference between the primary voltage andthe secondary voltage. The linear approximationcompensation reduces the voltage error without a trade-off
with the efficiency. However, the voltage error is notcompletely eliminated because the error is approximated aslinear. The pulse width neglected by the dead-time in the light

vac

vw

Vdc

N2

S2

Snt

V2

S4

PWM rectifier

010
001
V2

V2'

010
001

Sub
sector B

SectorVI

Sector I

Sub
sector A

100
V1
001
Vz : Zero vectors
001
Vz : Zero vectors
V5
V6
010
V7 100 V8 010 V9 001
100
010
001
Ex.) V1 100 Srp,Spr = ON, Ssp,Sps = OFF, Stp,Spt = OFF
001 Srn,Snr = OFF, Ssn,Sns = OFF, Stn,Snt = ON
※The number of ”1” means switch is turn on.
Sector V

Vz

Fig. 2. Fundamental voltage vector diagram to calculate the duty for thematrix converter.
Power transmission
Power transmission
Circulating mode
periodperiod
v1
v2
vpr 0
-v2
-v1
NVdc
vse 0
-NVdc

B

Iave

ipr 0

O

O
C’

T0 Ta Tz

(a)

A

C

T1

T2 Ta

T0

B’

Ta Tz T2

A’
Ta T0

T1

One switching cycle of the ripple cancel method
I2
2T2

i2
Iav e
i0

A
A

0

I1
2T1

Error I2

B
B

i2B
Iav e

Error I1

i0

I1

I2

T1

T2

I2
B
2T2

I1
2T1

A

A

0

I1

I2

T1

T2

(b) Conventional method(c) Ripple cancellation
Fig. 3. Ripple cancellation method in order to simplify duty calculation andminimize reactive power.

3 / 2

ib

Maximam line voltage v1
Midium line voltage v2

Da

I1

Db

I2

D0

DC detection voltage vout
Inductance value L, switching frequency fsw, Switching cycle Tsw

Fig.4 Open-looped control diagram with ripple cancelation method.

D1
D2
Carrier,area

Carrier Comparisonia

Duty Calculation
Ref.[5]Detectionline voltage

Line voltage
magnitude
relationship

S3

vse

N1

a

SectorIV

Voltage vector, 

vuv
vvw

L

Stn

Ssn
Sns

vb

Clark transformation

PLL

Srn

S1

SectorV2 Sector I
Vin* 100
III
V1 001
V1'
Vz
V1
v

001
V
100 4

Ideal value

ipr

Sector II

Fig. 4 shows the control block diagram with the open-loop
ripple cancellation method. In the ripple cancellation method,

iw

Stp
vpr

010
V
100 3

Fig. 3 shows the medium-frequency voltage and thecurrent waveforms that are the output of the matrix converter
and the secondary of the medium-frequency transformer
during the boost mode ( NVdc  2vac ) using the ripple
cancellation method. In the ripple cancellation method, theamplitude of the current vector during the power transmission
period is equaled to the medium-frequency current command
value Iave* by changing the output order of negative voltage
vector. As a result, the ripple current is equivalently neglected
in one switching cycle. The input current distortion is reduced
because the effect of the neglecting ripple current. The ripple
cancellation method is operated in both DCM (T0 > 0 and Ta= 0) and CCM (T0 = 0 and Ta > 0).

iu
iv

Spt

Ssp

Fig. 1. Circuit configuration of isolated AC-DC converter based on DAB
converter and matrix converter. The matrix converter directly generates themedium-frequency AC from the low frequency AC for DAB converter.

Fig. 2 shows the fundamental voltage vector diagram andthe current vector Iin*. In the matrix converter, the input
current outputs as a sinusoidal shape, SVM is used to calculate
a duty ratio for a converter. SVM is assumed that average
output current in each switching cycle is constant. The currnts
vectors I1 and I2 are close to current command vector. The
input current is controlled to follow the input command Iin* byadjusting the duty ratio of V1 and V2 in first half of cycle, andV4 and V5 in second half of cycle. However, the large riple
current occurs in the transformer current due to a very smalltransformer leakage inductance in one switching cycle, This
current variation, which is so-called "current ripple" in thispaper, causes the distortion of the input current and thetransmission power error.

Current
commandSps

Srp

3f -1f matrix converter

Fig. 1 shows the circuit configuration of a bidirectional
isolated AC-DC converter. The matrix converter directly
converts from the grid frequency in the medium-frequency
AC. A DC-link capacitor is not necessarry to the mediumfrequency isolated AC-DC converter based on the DAB
converter and the matirix converter. Furthermore, the powertransmission is realized only with an small inductor L whichis connected to the medium-frequency transformer. This
circuit does not use the large inductors and a DC inductor. As
a result, it is possible to reduce the system volume.

1
iu = I peak sin( −  )65iv = I peak sin( −  )61
iw = I peak sin( +  )
2

Spr

Snr

II. CIRCUIT CONFIGRATION AND CONTROL METHOD

I peak

vu iu
vv iv
vw iw

Input filter

load region. For these reason, the input current distortion is
not compensated completely. In order to improve thedistortion, the DC current feedback method is demonstrated toachieve constant instantaneous power of the input/output
power. Note that the input current distortion is reduced when
the instantaneous power factor is constant. The simulation
results show the usefulness of the DC current feedback
method.

Sru
Ssu
Stu
Srv
Ssv
Stv
Sap
San
Sbp
Sbn

the ideal three-phase current values iu, iv, and iw are obtainedusing the current command value Ipeak and the system phase θ.
The vector magnitudes I1 and I2 which are used in SVM arecalculated by α-β transformation and Clark transformation for
the three-phase current. The calculation of the duty ratio isperformed with the formula in Ref. [5].
III. VOLTAGE ERROR CAUSED DUE TO FOUR-STEP VOLTAGE
COMMUTATION

The matrix converters need the commutation thatprevents short circuits in a power supply and does not open
the inductive current pass. The commutation method isseparated into a voltage-based commutation and a currentbased commutation while satisfying these conditions. In thispaper, a four-step voltage-based commutation that does not
require a current sensor is applied [10].
Fig. 5 shows the model and the switching sequence of the
four-step voltage-based commutation using two bidirectionalswitches, two voltage sources, and the current source. Whenthe voltage-based commutation is made from the power
supply va to vb with the positive load current (iload > 0), thechanging of the current path from va to vb occurs when S1a isturned-off in the second step. In other words, the output
voltage is delayed by the dead-time Td from ideal timing.
Similarly, the commutation from va to vb occurs in the third
step under the condition of va < vb. The change of the actual
output voltage is delayed by 2Td. Therefore, the delay of theoutput voltage due to the voltage-based commutation is notalways same and is determined by the voltage condition andthe current direction. The compensation of the voltage error isrequired considering the delay.
A. Voltage commutation compensation during DCM
operation under boost condition
Fig. 6 shows the operation waveforms of the voltage-based
commutation during the DCM under the boost mode. The
relationship of the voltage magnitude is vu > vv > vw. The
voltage error of the matrix converter by the voltage-based
commutation is determined by the voltage magnitude and thecurrent direction. The two-level compensation using twocompensation values is used for the dead-time compensation
as a feed forward method. The two-level compensation
eliminates the voltage error caused by the dead-time [5]. The
actual pulse width differs from the ideal pulse width by thedead-time. The voltage error is eliminated by adding the deadtime compensation to the duty. The voltage output periods T2
and Tb change by the dead time Td due to the voltage
commutation. The period T2 becomes longer by Td, while theperiod Tb becomes shorter by Td. Therefore, the actual output
periods T2' and Tb' are expressed as (1) in the DCM operation.

Tb' = Tb + Td......................................................(1) 'T2 = T2 − Td
However, the overcompensation occurs by the two-levelcompensation near zero current, which results in the input
current distortion.
B. Voltage commutation compensation from DCM to CCM
operation
Fig. 7 shows the linear approximation compensation
method. The operation has the phase difference  between
the primary and the secondary voltages. The phase difference is affected by the voltage error when the operation mode

1

S1a Negative

va

S1b
S2b

vb

2

3

4

S1a
S1b
va > vb S2a
S2b

va
vb

or
Positive

Td

error

S1a
S1b
va < vb S2a
S2b

iload

S2a

va
vb

Td

error

Fig. 5. Commutation model and mechanism of voltage error.
Power transmission
Power transmission
period Circulating mode period
T0 Tb T1
(V9) (V1) (V1)

T2
(V2)

T0
(V9)

Tb T2
T1
(V4) (V5) (V4)

T0
(V9)

MC output voltage vpr [V]0
PWM rectifier
output voltage vse [V] 0

iave*
0Transfer primary current ipr [A](a) Ideal commutation
Power transmission
Power transmission
period Circulating mode period
T0 Tb T1
(V9) (V1) (V1)0
MC output voltage vpr [V]

Td

T2
(V2)

T0
(V9)

Tb T2
T1
(V4) (V5) (V4)

Td

T0
(V9)

Td

0
PWM rectifier
output voltage vse [V]Offset occurs

iave(iave < iave*)0
Transfer primary current ipr [A]Commutation delay

2Td 0

Td 2Td2Td Td2Td

2Td(b) Voltage commutation
Fig. 6. Middle frequency waveformsunder DCM.

Td

Voltage error

vpr 0

Td

Td

vpr 0



vse 0

vse 0

Tb

(Ta

0)

vpr 0


Ta Tb
(Ta < Td)

vse 0


Ta Tb
(Ta > Td)

Fig. 7. Linear approximation compensation method.

switches from DCM to CCM. The voltage error by the phasedifference  between the primary and the secondary voltages
occurs when the operating mode is switched from DCM to
CCM. The amount of the voltage error is nonlinear when the

idc

PI

Feedback control

K

PLL

iu
iv

1
iu = I peak sin( −  )65iv = I peak sin( −  )6

 i =Iw

iw

1
)peak sin( +2ia3 / 2

ib

Voltage vector, Detectionline voltage

vuv
vvw
vw

Line voltage
magnitude
relationshipMaximam line voltage v1
Midium line voltage

v2

Da

I1

Db

I2

D0
D1
D2

DC detection voltage vout

Sru
Ssu
Stu
Srv
Ssv
Stv
Sap
San
Sbp
Sbn

Carrier Comparison-

I peakDuty Calculation
Ref.[5]

I dc* +

Ideal value

iout

Clark transformation

DC current
commandCarrier,area

Inductance value L, switching frequency fsw, Switching cycle Tsw

Fig. 8. Feedback loop control diagram with ripple cancelation method.

commutation begins near zero current [11]-[12]. The current
direction at the start of the commutation changes due to theapplication of the two-level compensation. For this reason,
the two-level compensation causes the overcompensate. The
phase difference is greatly affected by the transmission powerin the circuit configuration of isolated AC-DC converter
based on the DAB converter and the matrix converter. The
error of the voltage output period based on the voltage error
seems to change the phase of the primary side and thesecondary side in the operation of the DAB converter. In
addition, the phase difference of the DAB converter causes alarge transmission power error even if the voltage error issmall. In a voltage-source PWM rectifiers, the compensation
performance is improved by adjustment of the thresholdcurrent [13]. However, the threshold current always changes
in the matrix converter because the instantaneous voltage
fluctuates at six times of the grid frequency. Due to this, thelinear compensation is applied for the phase difference
 under following three conditions. 1) Boundary conditions
with DCM (Ta => 0), 2) The duty ratio Ta is less than deadtime (Ta < Td), 3) The duty ratio Ta is longer than dead-time(Ta > Td). The linear approximation compensation amount Δ
is expressed as(Ta  0) Td = Td − Ta (0  Ta  Td )(2)
0(Ta  Td )

In chapter Ⅲ-A,B, we examined the two-levelcompensation method and the linear approximation method
in order to reduce the voltage error that occurs in the DCM orchanged operation mode from DCM to CCM during thecommutation. However, the voltage error is difficult tocompensate for the voltage output period shorter than thedead time. In addition, the linear approximation method does
not sufficiently reduce the voltage error. Thus, the input
current distortion is remind by these reasons.
C. Voltage commutation compensation with DC current
feed-back control
This chapter describes the problem of the linear
approximation, which is the problem of the feedforward
compensation, and the DC current feedback compensation,
which solves the problem of not being able to compensate theoutput of the pulses below the dead time. The instantaneous
power of the input/output power becomes constant due to the
DC current feedback. When the instantaneous power isconstant, the input current distortion does not occur under the
power factor is constant. Due to this, the problem of the

MC
(a)

Iout

L

PWM
ic
rectifier

vc

I

vdc

- '
I dc

+

Vdc

Circuit configuration with DC side.

I dc
*
dc

Idc

vL +
PI
control

+

vc
vc

- vc'

PI
control

iout

ACR
AVR
(b) Control diagram.
Fig. 9. Circuit configuration and Control diagramwith DC current feedback.

voltage error is solved without the compensation by thefeedforward.
Fig. 8 shows a block diagram with the feedback control.
In Fig. 4, the maximum input current value Ipeak is thecommand value, whereas in the feedback control, the DC
current Idc is used as the command. The current peak value iscalculated by a PI control and the output of the conversion
gain. The duty ratio is calculated by the equation in Ref. [5].
Fig. 9 shows the circuit configuration and the control
block diagram for the DC current feedback control. The aimof this compensation method is to keep the instantaneous
power of the input and the output power constant with thedirect current feedback. The input current distortion will
disappear under the constant instantaneous power and unity
power factor.
Fig. 9(a) shows the circuit configuration on the DC side.
The DC side circuit configuration has a LC filter between the
DC voltage source (battery side) and the PWM rectifier.
Fig. 9(b) shows the control block diagram. The automaticcurrent regulator (ACR) and the automatic voltage regulator(AVR) are applied to control the DC current. The PWM
rectifier output current is summation of the DC current Idc andthe capacitor current Ic. Furthermore, the capacitor current Ic
is zero in the steady-state. Thus, the PWM rectifier output
current Iout and the DC current Idc are equal in the steady state(Iout≒ Idc). In addition, the conversion gain K shown in Eq.
(3) is provided to convert the PI output value Iout to the
maximum input current value Ipeak.
^K=

I peak^I dc............................................................ (3)^Table 1. Experimental parameters.

^

I peak is the maximum value of the input current and I dc
is the output current at the rated operation. Note that these
conditions are considered in the steady state without theconversion losses. The configuration from the output of theconversion gain to the duty calculation is the same as in the
open-loop.
IV. EXPERIMENTAL VERIFICATION
Table 1 shows the experimental conditions. Note that the
rated power is 6 kW, the switching frequency is 50 kHz, andthe four-step voltage-based commutation is applied toprevent the commutation failures such as short-circuiting of
the input voltage. One-step cycle of the four-step voltagebased commutation is set to 250 ns.
Fig. 10(a) shows the primary side voltage, current, and thesecondary side voltage in the DCM region without thecompensation. The transmission power command is 2.0 kW.
From the experimental waveform, the voltage-based
commutation operation of the matrix converter causes thevoltage error in the output voltage because the transformer
current has the current offset during the zero-voltage period.
Fig. 10(b) shows the primary side voltage, current, andthe secondary side voltage with the two-level compensation.
The transformer current was zero in synchronization with thezero-voltage period by suppressing the output voltage error
by the two-level compensation.
Fig. 11 shows the primary-side voltage, current, and thesecondary-side voltage without the compensation at thetransmit power command is 2.0 kW. The input current
waveform is greatly distorted. This is caused by the nonlinear
voltage error in the phase difference  between the primary
and the secondary voltages when the operating mode isswitched from DCM to CCM.
Fig. 12 shows the primary-side voltage, current, and thesecondary-side voltage with linear approximationcompensation. The input current distortion is reduced byadding the error compensation based on Eq. (2) to the phasedifference .
Fig. 13 shows the input/output voltage and the current
waveform of the matrix converter and the rectifier at the rated
conditions under the boost ratio of 1.0. The input voltage andthe current are in the phase and controlled at THD of 5.0% orless at the power factor of 1.0.
V. SIMLATION VERIFICATION
Fig. 14(a) shows the simulation results without thevoltage error compensation. From the simulation waveform,a large input current distortion occurs due to the influence ofthe non-linear voltage error on the phase difference between
the primary and the secondary side voltage when theoperation mode is switched from DCM to CCM. The phaseerror between the primary and the secondary side occurs
during the voltage-based commutation. In addition, the large
input current distortion occure due to the large difference ofthe transmission power between DCM and CCM.
Fig. 14(b) shows the simulation waveforms with thefeedforward compensations that uses the two-levelcompensation and the linear approximation compensation. In
addition, the transmission power error during CCM is
improved and the distortion of the input current is reduced by

Quantity
Rated power
Three-phase AC voltage
DC voltage
Input frequency
Carrier frequency
Leakage inductanceTurn ratio of transformer

Symbol
P
vac
Vdc
f
fsw
L
N1:N2
Lf
Cf
Td

Input filter
Dead-time
vpr[500V/div]Value
6.0 kW
200 V
Max 100 V
50 Hz
50 kHz21 µH
3.3:1
0.19 mH
10 mF
250 ns4[msec/div]

0

4[msec/div]

vpr[500V/div]0
vse[100V/div]

vse[100V/div]00ipr[25A/div]

ipr[25A/div]00Offset is reduced

Offset is occured(a)Without compensation (b) With compensation
Fig. 10. Waveforms of matrix converter and rectifier.
Expanded waveform

iu[10A/div]CCM DCM

iu[10A/div]

0

0

0

0

0

0

vpr [200V/div] vse[50V/div]iL[20A/div]
10[msec/div]

10[msec/div]

Fig. 11. Operation waveforms without compensation.
Expanded waveform
CCM DCMiu[10A/div]

iu[10A/div]

0

0

0

0

0

0

vpr [200V/div] vse[50V/div]

Add Δ

iL[20A/div]

10[msec/div]

10[msec/div]

Fig. 12. Operation waveforms with linear approximation compensation.
0

vu[250V/div]0iu[20A/div]

Vdc[50V/div]0
idc[50A/div]

0

4[msec/div]

Fig.13. Input and output waveforms of matrix converter with rated
operation.

adding the error correction shown in Eq. (2) to the phasedifference.

Input voltage[V]200Fig. 14(c), it is demonstrated that the input current THD
is improved by the feedback compensation. However, thecompensation is not enough because of low feedback gain.
The optimization of the feedback gain will be revealed withstability analysis in future works.

0-200

Input current [A]CCM DCM

100Fig. 15 shows the comparison of the input current THDs
among each compensation method. The input current THD isredeced by 70% due to applying the feedforward
compensation. Howevere, more improvement is required lessthan 0.4 p.u. due to the insufficient compensation of the linear
approximation method In the simulation results, it seems thatthe DC current feedback compensation is better than thefeedforward compensation at 0.2 p.u. or more. However, thegain of the PI controller is set to very high-response in thesimulation. The optimization of the controller, which applies
the feedback compensation in parallel to the feedforward
compensation, is needed in future works.

-10
20 ms
(a)
200

-200

Input current [A]CCM DCM

10

-10
20 ms

(b) With feedforward compensation(Two-level compensation + linear approximation compensation)
Grid voltage[V]
200
0
-200

Grid current [A]

10

[3]

[4]

[5]

[6]

[7]

THD:4.0 %020 ms(c) Feedback compensation
Fig. 14. Simulation result of the operation mode switchingfrom DCM to CCM.
20
20
20

current
Input
Grid
currentTHD
THD [%][2]

THD:6.8 %0

REFERENCES
Muhammad Hazarul Azmeer bin Ab Malek, Hiroaki Kakigano, andKiyotsugu Takaba"Combined PulseWidth Modulation of Dual Active
Bridge DC-DC Converter to Increase the Efficiency of Bidirectional
PowerTransfer,"IEEJ Journal of Industry Applications, vol. 7, no. 2, pp.
166-174, (2018).
Ryota Kondo, Yusuke Higaki, Masaki Yamada, “Experimental
Verification of Reducing Power Loss under Light Load Condition of a
Bi-Directional Isolated DC/DC Converter for a Battery ChargerDischarger of Electric Vehicle”, IEEJ Journal of Industry Applications,
Volume 10, Issue 3, pp. 377--383, 2021
J. Afsharian, D. David Xu, B. Gong and Z. Yang, "Space vector
demonstration and analysis of zero-voltage switching transitions inthree- phase isolated PWM rectifier," 2015 IEEE Energy Conversion
Congress and Exposition (ECCE), Montreal, QC, 2015, pp. 2477- 2484.
S. Takuma, K. Kusaka, J. Itoh, Y. Ohnuma, S. Miyawaki: "A Novel
Current Ripple Cancellation PWM for Isolated Three-phase Matrix
DAB AC-DC Matrix Converter", EPE2019, Vol. , No. , pp. (2019)
J. Itoh, S. Nakamura, S. Takuma, H. Watanabe: "Isolated Three-phase
AC to DC converter with Matrix Converter Applying Wide Output
Voltage Operation", ECCE-US (2020)
Lukas Schrittwieser, Michael Leibl, Johann W. Kolar, R. Baranwal and
N. Mohan, "99% Efficient Isolated Three-Phase Matrix-Type DAB
Buck-Boost PFC Rectifier" in IEEE Transactions on Power Electronics(2019).
Shunsuke Takuma and Jun-ichi Itoh, “Two-step Commutation for
Three-phase-to-single-phase Matrix Converter with High-frequency
Isolation,” IEEJ J. Industry Applications, vol. 8, no. 4, pp. 704–712,(2019).

Without compensation
Input voltage[V]0VI. CONCLUSIONSIn this paper, the compensation method of the voltage
error was proposed utilizing the voltage-based commutationof the matrix converter. The voltage error caused by thevoltage-based commutation was investigated by using thetwo-level compensation and the linear approximationcompensation methods. The experimental results showed thatthe compensations were effective in reducing the input
current distortion. THD of the input current was improved by
up to 70%. However, the voltage error compensation was notcompletely reduced in light load due to the linear
approximation. Therefore, the DC current feedback
compensation was demonstrated by the simulation to reducethe influence of the voltage error and insensitive less than thedead time. The simulation results showed that the voltage
error was improved compared to the method using thefeedforward compensation.
[1]

THD:20.0 %Without compensation(Experimental reult)
補償なし
Feed-forward compensation(Experimental reult)
フィードフォワード補償
Feedback compensation(Simulation
result)フィードバック補償

15
15
15

10
10
10

55

000
0

0.2
1200

0.4
0.6
0.4
2400
3600
0.6
Output power Pout [p.u.]

0.8
0.8
4800

1.0
1.0
6000

Fig. 15. THD characteristics (boost ratio 1.0).

[8]

[9]

Prathamesh Pravin Deshpande, Amit Kumar Singh, Hau Chong Aih,and Sanjib Kumar Panda, “A Matrix-based Isolated Bidirectional ACDC Converter with LCL type Input Filter for Energy Storage
Applications,” IEEJ J. Industry Applications, vol. 8, no. 4, pp. 644–
651, (2019)
Daiki Yutaka, Masato Koyama, Naoki Yamamura, Muneaki Ishida,"Proposal for Universal Direct Converter Based on Matrix Converter
for Three-phase AC and DC Source", IEEJ J. Industry Applications,
vol.9, no.4,pp.428--434, (2020)

[10] Jochen Mahlein, Jens Igney, Jörg Weigold, Michael Braun, and Olaf
Simon “Matrix Converter Commutation Strategies With and Without
Explicit Input Voltage Sign Measurement” IEEE Transactions on
Power Electronics, Volume: 49, Issue: 2, Jan. (2002).
[11] Seon-Hwan Hwang, Jang-Mok Kim, “Dead Time Compensation
Method for Voltage-Fed PWM Inverter” IEEE Transactions on Energy
Conversion, Volume: 25, No: 1, Mar. (2010).

[12] K. Takagi and H. Fujita, "Dynamic control and dead-timecompensation method of an isolated dual-active-bridge DC-DC
converter," 2015 17th European Conference on Power Electronics andApplications (EPE'15 ECCE-Europe), Geneva, 2015, pp. 1-10
[13] Tomoyuki Mannen and Hideaki Fujita “Dead Time Compensation
Method Based on Current Ripple Estimation” IEEE Energy
Conversion Congress and Exposition (2013)