電気学会論文誌●（●●●●●●●部門誌）
IEEJ Transactions on ●●●●●●●●●●●●●●●
Vol.●● No.● pp.●-●●
DOI: ●.●●/ieejeiss.●●.●

Paper

Isolated DC to Single-Phase AC Converterwith Active Power Decoupling Capability Using Coupled Inductor
Nagisa Takaoka＊ Student member
Hiroki Watanabe＊ Member,
Jun-ichi Itoh＊a) Senior Member(Manuscript received Jan. 00, 20XX, revised May 00, 20XX)
This paper proposes a novel active power decoupling circuit that is integrated with an interleaved boost converter for isolated
DC to single-phase AC conversion applications. The proposed isolated converter consists of two full-bridge inverters with a coupledinductor, a small buffer capacitor for active power decoupling, and a diode rectifier. Further, the proposed isolated converter iscontrolled for active power decoupling and DC to single-phase AC power conversion independently by the coupling inductor andcommon-mode operation of the interleaved boost converter, respectively. This paper presents the control method of the proposed
converter and design criteria of the passive components. In addition, the performance of the proposed isolated converter isdemonstrated by the experimental results in order to confirm the validity of the proposed method. As the experimental results, thesecond-order harmonic due to the double-line frequency power ripple is reduced by 84.5% owing to the active power decoupling
capability. Finally, a maximum efficiency of 94.5% is obtained using the proposed converter.
Keywords : Isolated DC to single-phase AC converter, Power decoupling, Coupled inductor;

1.

order to solve this problem [8-15]. In these methods, the doubleline frequency power ripple is compensated by the small capacitor
which permit the large voltage fluctuation. As a result, the small
capacitor such as firm or ceramic capacitor is chosen for the DClink capacitor and the active power decoupling capacitor. However,almost conventional active power decoupling circuits require theadditional components including the passive components and theswitching devices for the active power decoupling circuit and also
decrease the power density of the isolated converter. Therefore, theactive power decoupling method with few additional components isrequired in order to achieve the high-power density converter [14].
This paper proposes a novel isolated DC to single-phase AC
converter with active power decoupling capability. This coupledinductor is integrated into the boost chopper of the interleaved
converter and the active power decoupling circuit by setting thethird winding to the coupled inductor. This integrated circuit using
the coupled inductor is achieved to control the common-voltage as
an active power decoupling operation mode and the differentialvoltage as an operation mode of a power conversion with thegalvanic isolation independently. Therefore, the proposed converter
greatly reduces the number of the switching devices and additional
components for the active power decoupling circuit. In addition, thesecondary side of the proposed converter reduces the switchinglosses because the secondary side converter is operated at the twice
grid frequency to only control the polarity conversion. Thus, theproposed converter can be expected to achieve the high-power
density and the long-lifetime. It is noted that the integrated topology
into the active power decoupling capability and the power
conversion have been researched in the past work [16-19]. These
active power decoupling methods do not require additionalswitching devices by applying common and differential operation
modes to the interleave topology. However, these active power
decoupling methods have problems that there are plural additional

Introduction

Recently, a Fuel Cell (FC) has been widely accepted as therenewable energy sources for the energy saving [1-2]. The power
conditioning systems (PCSs) are the one of the applications for FC,
and it may become a trend for next power generation systems assame as photovoltaic systems. PCSs require some capabilities: (1)boost-up function of the FC voltage, (2) galvanic isolation, and (3)power decoupling capability between DC and single-phase AC. In
order to satisfy these requirements, PCSs typically consist of an
isolated DC to single-phase AC converter which includes the boost
converter, isolated DC/DC converter, and a voltage source inverter.
In addition, a lot of circuit configurations for PCSs have beenstudied in order to achieve the high-power density. The interleaved
boost converter is one of good options for the reduction of thevolume of the boost inductor because the current ripple is cancelledby the phase-shifted operation regardless of the small inductor [36]. In addition, the coupled inductor shares the multi winding during
each phase, and the number of cores is reduced in comparison withthe multi-phase interleaved converter.
On the other hand, the double-line frequency power ripple is theone of the primitive issues for the single-phase application because
it fluctuates the DC power with the twice of the grid frequency.
Therefore, a bulky electrolytic capacitor is generally placed on theDC-link to absorb the power ripple. It is do called “passive power
decoupling”. However, the electrolytic capacitors may limit thelife-time, and increases the circuit volume [7].
In contrast, active power decoupling, which uses passive
components with active switches, circuits have been proposed ina) Correspondence to: Jun-ichi Itoh. Email:itoh@vos.nagaokaut.ac.jp＊
Nagaoka University of Technology
1603-1, Kamitomioka, Nagaoka, Niigata 940-2188, Japan© 200● The Institute of Electrical Engineers of Japan.

1Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)energy components for the active power decoupling circuit. In thesolution method of the elimination of the additional components forthe active power decoupling circuit, the isolated converter with thecenter-tapped transformer has been reported in [18]. This active
power decoupling topology is eliminated the additional switching
devices and consists only one buffer capacitor and one buffer
inductor, but that the required capacity of the energy buffer
components increases in the case of the low input DC voltage.
The originality of this active power decoupling topology is not
required for any additional components, owing to the employmentof the coupled inductor. In addition, the proposed method verifies
the active power decoupling capability in the low input DC voltage.
This active power decoupling method by using the coupled inductor
was also introduced in [21]. However, in this paper, only the basic
verification of the scale model is reported, and the detailed
designing method of the circuit condition and the coupled inductor
to realize the proposed control method and the active power
decoupling capability is not clarified. Thus, the circuit presented in[21] is not reproducible and unclear whether it can achieve a highefficiency.
This paper focuses on clarifying the designing flow and theverifying superiority of the proposed method based on the analysis
results of the conventional circuit in [21]. Furthermore, it was
confirmed by the experimental analysis that the semiconductor
losses are reduced compared to the conventional circuit with theactive power decoupling capability.
This paper is organized as follows; first, the proposed converter
for the DC to single-phase AC converter and the principle of theactive power decoupling method are introduced; second, theoperation mode of the proposed converter and the modulation
methods with the active power decoupling capability and thedesigning flow of the coupled inductor are explained; finally, thefundamental operation waveforms of the proposed converter are
evaluated in experiments.

2.

igrid
Lboost
Lload

Cdc

Lboost

Cdc 2

vgrid

vout

idc
Vdc
Interleaved
Isolated DC-DC
Full bridge
converter
converter
InverterFig. 1. Conventional isolated DC to single-phase AC converter withinterleaved boost topology. The conventional converter uses a bulky
electrolytic capacitor Cdc to absorb the power ripple component caused
by a single-phase ac load.
Active power decoupling circuit

igrid
Lboost
L1
Lboost

Lload
Cbuf

Cdc 2

vout

vgrid

idc
Vdc
Interleaved
Isolated DC-DC
Full bridge
converter
converter
InverterFig. 2. Conventional isolated DC to single-phase AC converter withactive power decoupling circuit. The conventional active power
decoupling circuit consists of few passive components and switching
devices.

S2a p S2bp

S1a p S1bp

iload

N3

Lboostidc
Vdc

Circuit Topology

2.1
Conventional Circuit
Figure 1 shows a conventional isolated DC to single-phase AC
converter. This converter consists of an interleaved boost converter,
an isolated DC to DC converter, and a full bridge inverter. Theinterleaved boost converter has the capability of the boost-up of the
input voltage. This two-phase interleaved topology is employed toreduce the conduction loss and the volume of the boost inductor. In
addition, the full bridge inverter at the primary side generates ahigh-frequency square voltage to reduce the volume of thetransformer. However, a bulky electrolytic capacitor Cdc2 istypically required on the DC-link in order to compensate thedouble-line frequency power ripple.
Figure 2 shows the isolated DC to single-phase AC converterwith the conventional active power decoupling circuit. The
conventional active power decoupling circuit in Fig. 2 consists of
two additional switching devices and two passive components. The
conventional active power decoupling circuit regulates the buffer
capacitor voltage, and the power ripple is absorbed by the small
capacitor of Cbuf. However, these additional components increase
the circuit volume and decrease the conversion efficiency. In
particular, the addition of the magnetic component constricts thehigh-power density of the converter system.

N1
N2

Vcom

vdif Cbuf

vs
vCbuf

vout

Lload
vload

S2a n S2bn
S1a n S1bn
: Integrated part for power decoupling capability

Fig. 3.
Proposed isolated DC to single-phase AC converter. The
primary-side converter together with the coupled inductor and thebuffer capacitor are controlled in order to convert the DC input power
to the AC output power and absorb the power ripple without additionalswitching devices and inductors.

2.2
Proposed Circuit
Figure 3 shows the proposed isolated DC to single-phase AC
converter with the active power decoupling capability. The
proposed converter consists of two full bridge inverters, the buffer
capacitor, the diode rectifier and the coupled inductor. The
proposed converter integrates with the active power decoupling
circuit, interleaved boost converter, and isolated DC/DC converterin order to reduce the number of components. The proposed method
is operated two operations as an active power decoupling and apower conversion at the same time by applying the common anddifferential voltage control with the coupled inductor. The active
power decoupling control is employed with a common mode
operation, whereas the grid current control is implemented with adifferential mode operation.
The secondary converter converts the high-frequency AC to thesingle-phase line frequency AC without DC-link. The secondary
converter performs the rectifier operation and the polarity reversion

2

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)

t

0

igrid

igrid= 2Igrid sin(ωot)

charge
t

0

t

0

t

0

Buffer powerpbuf [W]pgrid＝chargedischarge

Input power
Pdc [W]........................... (1),

＝

pbuf

0

pout = 2Vload sin (ot )  2 I load sin (ot )



＝

pdc

= Vload I load 1 − cos ( 2ot )tdischarge

3.1
Parameter design of Active Power Decoupling
Figure 4 shows the principle of the active power decoupling
method. It is assumed that firstly, the instantaneous single-phase AC
power pout is expressed as

＝
t

0

t

Output power
pgrid [W]Fig. 4. Relationship among each waveform.

= Pave − Pave cos ( 2ot )achieved, whose result is that the power ripple component at DCside is absorbed.
The buffer capacitor in the active power decoupling is calculatedbywhere Vload and Iload is the root-mean-square (rms) value of theoutput voltage and current, ωo is the grid side angular frequency andPave is the average output power. Note that a grid current igrid andvoltage vgrid are a sinusoidal waveform with a unity power factor.
In order to obtain a constant value of the input current at the DC
side, the ripple component should be eliminated. Thus, the buffer
power pbuf, which absorbs the power ripple component generated
by a single-phase AC load is assumed by the following:

Cbuf =

Pave............................................................... (6).
ovave vc

In the case with the active power decoupling method, the buffer
capacitor voltage is controlled to large Δvc. Then, the required
storage energy is achieved with small capacitance. This is theprinciple to be able to reduce the capacitance in the active power
decoupling circuits. It should be noted that the limited bottom valueof the buffer capacitance is decided by the voltage capacity of theswitching devices in the primary-side. In addition, the averagevoltage vave inevitably becomes the twice of the input voltage Vdc,
which the turn number N1 is equal to the turn number N2 of thecoupled inductor. The designing method of the turn numbers of thecoupled inductor is explained in the fourth section.

pbuf = Pave cos ( 2ot ) ........................................................ (2).
The polarity of the buffer power is defined to be positive when
the buffer capacitor is charged.
1
PC = oCbuf vCbuf 2
2................... (3),
2
2

1
v  
v  
= oCbuf  vave + c  −  vave − c  
22  
2  
where vave is the average voltage of the buffer capacitor and Δvc isthe peak-to-peak of the buffer capacitor voltage range. The buffer
capacitor voltage vCbuf which needs to absorb the power ripple iscalculated based on the energy of the buffer capacitor. First, theenergy of the buffer capacitor WCbuf is expressed by a voltagecurrent equation of a capacitor and (2).

3.2
Duty Calculation
Figure 5 shows the operation modes of the primary-side
converter. The proposed primary-side converter has four operation
modes which generate a phase difference between the switching
timing of each arm as an interleaved converter. The boost inductor
Lboost is a leakage inductor of the transformer, and it is utilized forthe boost operation and the current control for the active power
decoupling operation. Note that if the inductance value of theleakage inductor is insufficient for the designing value, a small
inductor may be added. The boost inductor is decided by the current
ripple ratio in the worst case, which is calculated by using the
maximum on-duty ratio of the common mode operation.

t
dvCbuf WCbuf =  vCbuf  Cbuf
 dt
t0
dt 

........................................ (4)=  Pave cos ( 2o ) dttt0

By using (4), the buffer capacitor voltage to compensate thepower ripple is presented by the following:vCbuf = VC 0 2 +tvgrid= 2Vgrid sin(ωot)

0

ibuf

0

vgrid

＋

Idc

Control Strategy



t

0

＋

3.

vCbuf
VCmax
VCmin

Vdc

＋

of the input voltage to output the grid voltage. Therefore, theswitching loss of the secondary full-bridge inverter is smaller byapplying the polarity reversion at 50 Hz. The snubber circuit, whichconnected between the diode rectifier and the full-bridge inverter,
has a role of an absorbing surge generated by the connection
between the load inductor, the leakage inductor and the boost
inductor. Note that the resistor in the RCD snubber circuit is todischarge the stored energy in the snubber capacitor. The proposed
converter realizes the DC to single-phase AC power conversion andthe active power decoupling capability with lower the number ofthe components in comparison with the configuration of Fig. 2.

Lboost =Pavesin ( 2ot ) ........................................ (5),oCbufwhere Vco is the buffer capacitor voltage at a start time t0. In the
proposed converter, the input current idc flowing to the coupledinductor from DC supply is controlled as a constant DC value. Thus,
the buffer capacitor voltage has the power ripple components. The
proposed converter with the active power decoupling capability isidc =

dcom _ muxVdc
2idc f sw

iripple _ pp / 2
idc............................................................ (7),.............................................................. (8)where Δidc is the allowable current ripple ratio, iripple_pp is the peak
to peak value of the input current and fsw is the switching frequency
3

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)of the interleaved converter. Note that the dcom_max is the maximum
reference of the common mode operation in one period, which is
explained in detail in the next section.
In the proposed system, the differential-mode voltage of thecoupled inductor is controlled to generate the output current,
whereas the common-mode voltage is controlled to perform theactive power decoupling capability. In the control of thedifferential-mode voltage, the three-winding voltage of the coupledinductor is controlled into a three-level voltage by operation modes
as shown in Fig. 5 (a) and (b).
On the other hand, the common-mode voltage vcom is controlled
to absorb the power ripple in the control of the common mode
voltage. Especially, the power ripple component occurs in the buffer
capacitor voltage, because the input current is regulated into a
constant value in the active power decoupling method of the
proposed converter. Moreover, the active power decoupling
capability consists of a boost topology in the proposed operation.
The coupled inductor is boosted the fluctuation of the buffer
capacitor voltage in bigger than the input voltage Vdc. The
capacitance of the buffer capacitor is achieved to reduce enough toincrease the fluctuated voltage of the buffer capacitor to absorb thepower ripple component. Thus, the proposed converter with theboost topology of the active power decoupling capability isachieved to reduce the capacitance than a buck-type topology of theone [14].
Each duties of the primary converter in one carrier period should
satisfy the following requirement:

ic
VCbuf

S1a p
vcom
Lboost S1a n

S1bp

Lboost

Cbuf
idc
Vdc
(c) Common mode operation 1

Fig. 5.
inductor.

Vdc

S1bp

S1a n

vdif VCbuf
S1bn
Cbuf

idc

ic

(b) Differential mode operation 2
S1a p

ic

vdif
VCbuf
S1bn

S1a p
vcom

Cbuf

idc

(a) Differential mode operation 1

vcom
Lboost S1a n

S1bp

ic

vdif VCbuf
S1bn
Cbuf

idc
Vdc
(d) Common mode operation 2

Operation modes of primary-side converter with coupled

1

Carrierdcom+ddif
dcom
0

A
B
S1ap
S1bp

1
0

1
0
1
0
1
0

Fig. 6.
Relationship between gate signal and each command inprimary converter.

second winding of the coupled inductor. Thus, the on duties of thecommon mode operations as shown in Fig. 5(c) and (d) areequalized to keep the average voltage of the buffer capacitor, so
maximum on duty of the common mode operation in one switching
period is divided equally to 0.5.
Figure 6 shows the relationship among each command, thetriangle carrier and the gate signals of the primary converter. In the
differential mode operation, the differential voltage pulses between
the first and second winding are controlled as PWM pulses. Then,the period of each duty as shown in Fig. 5 (a) and (b) in one carrier
period is needed to be equal in order to suppress a DC biased
magnetization of the transformer, especially. The comparing signals
A and B are generated by comprised duties which the sum of dcom
and ddif, only dcom and a triangle carrier as shown in Fig. 6.
Therefore, the desired operation in each duty of the primaryconverter is actualized as described previously.
Figure 7 shows the control block diagram of the proposed
converter with the active power decoupling capability. The current
controls and the voltage control are employed with PI controllers.
The input current is controlled to a constant DC value to absorb the
double-frequency power ripple component. This input current
control is applying as a minor loop of the buffer capacitor voltage
control. This buffer capacitor control controls only the average of

2Vload sin(ot ) ................................... (10)vCbuf

where vout* is a command of the output voltage as a grid-side.
In contrast, the duty of the common mode operation is expressedby using the calculation result from (10) in order to meet the control
condition (9).
v *d com = d com _ c = (1 − d dif ) com ......................................... (11)vCbuf

where vcom* is a voltage command of the common mode operation,
which is output command by the input current control as explainedin the next section. The maximum on duty ratio of the common
mode operation is around the zero-voltage period of the grid voltage
as shown in (11). The maximum on duty of the common mode
operation dcom_max, which has the condition of the worst case of thecurrent ripple ratio is presented bydcom _ max =Lboost S1a n

vdif
S1bn

Vdc

where ddif_a and ddif_b are the duties of the differential mode
operation as shown in Fig. 5 (a) and (b), whereas dcom_c and dcom_d
are the duties of the common mode operation as shown in Fig. 5(c)and (d). If the output voltage for the grid side is controlled into asinusoidal waveform, the voltage pulses between the first andsecond winding are controlled as PWM pulses. The sum duties ddifof the differential mode operation is given byd dif = d dif _ a + d dif _ bvout *
=
vCbuf

S1bp

vcom

1 = d dif _ a + d dif _ b + d com _ c + d com _ d ...................................... (9)

=

S1a p

vcom* vcom*................................................ (12).
=
VC 0
2Vdc

In the case of the zero-voltage period of the grid voltage, the buffer
capacitor voltage becomes the average voltage as a twice of DC
input voltage by using same turn numbers between the first and
4

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)

idc*
PI

2

Filteridc
Input current control

vCbuf
Average voltage
control for Cbuf
igrid*PI

Fbuf (s)

1
vCbuf

×

+
-

Pulse patternconversion

vCbuf_ave*

Normalization of buffer voltage reference

vcom*

1
Carrier

PI

vout*

1

ABS

vCbuf
igrid
Current control of output filter inductor

+

vdif*

polarity
conversion

S1ap
S1an
S1bp
S1bn
S2ap
S2an
S2bp
S2bn

Fig. 7. Proposed control block diagram. The DC current is controlled to absorb power ripple. The buffer capacitor voltage control is employed in order toavoid the divergence of the average voltage due to discretization.

the buffer capacitor voltage vCbuf_ave in order to avoid the divergence.
The average value of the buffer capacitor voltage vCbuf_ave is twiceof the input voltage Vdc. In the proposed converter, the output
current is controlled by the full bridge inverter at the primary side.
Then, the secondary converter has the role as the rectifier and thepolarity conversion.

4.

I pV p + I 3V3

Ku
J
iload

Selection of core
(Output: Ae, W, Bs)

Vdc
vcbuf
vout

The coupled inductor of the proposed converter used for agalvanic isolation at the power conversion and the current control
for the active power decoupling operation and the power conversion
to single-phase AC side. The designing method and each turnnumber of the coupled inductor in the proposed converter is
explained in this section.
Figure 8 shows the flowchart of the designing method for the
coupled inductor in the proposed converter. The designing
condition as input elements for the starting flowchart is that fsw isthe switching frequency of the interleaved converter, Ku is awindow utilization factor and J is a current density of a winding.
The voltage and current values of other input elements are setup torefer an application of the converter system. It is noted that the
coupled inductor of the proposed converter definitely is made byusing only one core in order to cancel each magnetic effect.
First, these parameters of the core are decided by used to refer
the switching frequency, which is selected by the semiconductor
devices and the rated power of the converter system. In the
proposed converter, the switching frequency of the interleaved
converter is decided on 50 kHz by applying the SiC semiconductor
devices in this paper. The parameters as output elements of theselecting core are that Ae is a cross-sectional area, W is a windingarea and Bs is a saturated magnetic flux of the core. These
parameters have an influence on the selected core size.
Second, the coupled inductor with the third winding is designed
by focus on the maximum magnetic flux density. In general, themagnetic flux density Bac is expressed by using the area productmethod in [20],WAe f sw JK u K f

Start

fsw

Designing Method of Coupled Inductor

Bac =<Input>Calculation of maximum
on time tdif by (14)-(16)Design of maximum
magnetic flux
density Bmax by using (17)Bs>Bmax ?

noyesCalculation ofturn number N by (18)-(19)Ku < 1 ?

no

yes

End
Fig. 8.

Flowchart of designing method for coupled inductor.

a coupled inductor [20]. From (13), the maximum magnetic flux
density is calculated by using a constant input voltage and aconstant on-time in the switching period in general.
In the proposed operation, the input and the third winding voltage
as a differential voltage vdif are not a constant value but expressedby the PWM pulses with the twice of the grid frequency. Thus, themaximum magnetic flux density in the proposed operation depends
on the twice of the grid frequency. The coupled inductor of the
proposed converter requires the deriving condition; (1) differential
voltage at the maximum point of the magnetic density, (2)maximum on-time of the differential mode operation.
Figure 9 shows each voltage command and the integrated
waveform of the differential voltage in the proposed operation. The
duties of the two operation modes as shown in Fig.9 (a) are distorted
because they are calculated by using the square root of the twice ofthe grid frequency and the grid frequency in order to control with anormalization. On the other hand, Fig.9 (b) shows the voltage
waveforms of the common and the differential mode operation, andFig.9 (c) shows the expansion in Fig.9 (b). The differential-mode
voltage is a three-level voltage waveform, which is combined ofvCbuf, -vCbuf, and a zero-voltage period. The period of the zero-..................................................... (13)where Ip is an input current as a sum of the first and second winding
current, Vp is an input voltage as a sum of the first and second
winding voltage, I3 is a third winding current and V3 is a thirdwinding voltage. In addition, Kf is a waveform coefficient, which is
decided by the kind of input waveform with square or sine wave for
5

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)Buffer capacitor voltage vCbuf

Duty of differential mode operation ddif

Max.

2Vdc

Command of output voltage vout *0Common mode voltage vcom

Duty of differential mode operation ddif

Max.

Common mode voltage vcom

vCbuf
0.5vCbuf

0

Duty of common mode operation dcom
0.5

0

0

Differential mode voltage vdif00

-vCbufDuty of differential mode operation ddif Integrated value of vdif

0.0

5.0

10.0

15.0

[ms]

0.0

Integrated value of vdif000Differential mode voltage vdif

vCbuf

5.0

10.0

[ms]

15.0

5.58

5.60

5.62

5.64

5.66

[ms]

(a) Each command waveform(b) Integrated value of differential voltage(c)Expansion of each waveform
Fig. 9.
Relationship of differential voltage as input voltage for coupled inductor and integrated value of differential voltage. The integrated value ofthe differential mode voltage is fluctuated with the grid frequency because the gate signals of the differential mode is generated by PWM pulses as a gridsine waveform.

voltage level in the differential-mode voltage is utilized with the top
and the bottom level of the common-mode voltage. From Fig.9 (b),the integrated value of the differential-mode voltage as a magnetic
flux has an envelope curve caused by PWM operation and the twiceof the grid frequency. Therefore, the magnetic flux density, which
is proportional to the magnetic flux has the same envelope curve
and the maximum value at the same phase as the maximum pointof the duty ddif in the proposed operation.
In order to derive the maximum magnetic flux density Bmax in the
proposed operation, it is needed to define the on-time of thedifferential mode operation, which is calculated by using theswitching frequency and the duty of the differential mode operation
ddif. This on-time of the differential mode operation is used toreplace from the waveform coefficient in (13) to the calculation
condition of the proposed operation. The on-time of the differential
mode operation tdif is expressed by
tdif =

d dif
f sw

=

Bmax =

WAe JKu.............................. (17).

3.27 I loadVload

WAe JKu f sw

Note that maximum magnetic flux density is calculated with theconditions, which are the same turn numbers between the input-side
and third-winding side, and excluding the core loss. If the obtained
result of the maximum magnetic flux density is smaller than thesaturation magnetic flux density of the selected core, the designing
condition is satisfied.
Finally, each turn number is calculated by using each voltage
condition and the maximum magnetic flux density. The important
point of the design for the coupled inductor is the turn ratio of the
input-side, which consists the number of turns at the first-winding
N1 and the second-winding N2 as shown in Fig. 3. The turn ratio N1
vs N2 makes sure to set up 1:1, because the effect of the magnetic
flux at each winding are canceled each other by the same numberof turns. Thus, the designing method is as same as a generaltransformer designing method regardless of the ripple component
as a 50 Hz. The turn number is presented by

2Vload sin(ot )................ (14).
P
VC 0 + ave sin ( 2ot )  f sw
oCbuf2

The angular frequency ωo’ at the maximum on-time of thedifferential operation mode is given by

N = 0.5N1 = 0.5N2 = N3 .............................................. (18),16
arg max tdif (ot )   t = o 't ..................................... (15).
25N=

As described above, in the proposed operation, PWM
operation for the grid current control is performed by using thenormalization with the twice of the grid frequency. The
maximum magnetic flux density deviates from the angularfrequency at the maximum point of the on-time. The maximum
on-time tdif(ωo’) of the differential mode operation is represented
by using (14)-(15) and (5),tdif (o ' ) 

2iload (o ' )vCbuf (o ' )tdif (o ' )vCbuf (o ' )tdif (o ' )2 Bmax Ae

0.45 2Vload............................ (19)
Bmax Ae f sw

where N3 is the third-winding of the coupled inductor. For
simplicity and voltage conditions, the turn ratio N1+N2 vs N3 sets up
1:1 in this paper. The turn ratio is freely decided by the relationshipwith the input and the output voltage.
It is necessary to confirm whether the window utilization factor
Ku exceeds 1.0 from the derived turn number and the winding areaof the selected core W. If the window utilization factor is less than
1.0, the design of the coupled inductor with the three winding ends.

0.91 2Vload........................... (16).
P
VC 0 2 − 0.77 ave  f sw
oCbuf

5.

The maximum magnetic flux density in the proposed operationis represented as following;Experimental Results

Table 1 shows the experimental conditions with R-L load, andFigure 10 demonstrates the experimental waveforms of the
proposed converter. It is noted that the unit capacitance constant inTable 1 is defined as the electrostatic energy of the capacitor divided

6

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)Table 1

Experimental condition

Input voltage
Rated power
Boost Inductor (Lboost)

Load voltage
200 Vdc
Load frequency
2 kW
Filter Inductor (Lout)
1.1 mH
Buffer Capacitor (Cbuf) 200 mF(8 ms) Turn ratio of coupledinductor N1:N2:N3
Load current
10 Arms
Carrier frequency of
Switching frequency of50
kHzfull bridge inverter
secondary-inverter

Buffer capacitor voltage vCbuf 50V/div.

200 Vrms
50 Hz
2.0 mH(3.1%)

Input current idc 5A/div.

1: 1:250 Hz

0Buffer capacitor voltage vCbuf 100V/div.

00

DC input voltage Vdc 100V/div.

0
vCbuf

Common mode voltage vcom 200V/div.

Output current iload 5A/div.
Output voltage vout 200V/div.
(a)0

4 ms/div.

Without active power decoupling method

0vcom

Buffer capacitor voltage vCbuf 50V/div.
Input current idc 5A/div.

0

vdif

Differential mode voltage vdif 500V/div.

8 µs/div.

Fig. 10.
Extension of each voltage waveforms. The maximum
voltages of the two operation modes are decided by the buffer capacitor
voltage with the twice grid frequency.

0

Output current iload 5A/div.

00

by the rated power of the converter, and its unit is “second” [22].
From the experimental waveforms with the coupled inductor, thefundamental operation of the proposed method was confirmed thatthe differential-mode voltage vdif is a three-level voltage waveform
based on a zero voltage. On the other hand, the common-mode
voltage vcom is a three-level voltage based on a half of the buffer
capacitor voltage. Therefore, the proposed converter greatly
reduces the number of the switching devices and additional
components for the active power decoupling circuit, owing to theemployment of the coupled inductor.
Figure 11 shows the experimental results with and without the
proposed active power decoupling method. In the experimental
result without the active power decoupling operation as shown in
Fig. 11 (a), the capacitance of the buffer capacitor Cbuf is not enoughto absorb the double-line frequency component on the dc sidecompared by using the electrolytic capacitors. As a result, theremaining double-line frequency component occurs to the dc input
current. On the other hand, from Fig. 11 (b), the buffer capacitor
voltage is operated to absorb the power ripple component.
Therefore, the double-line frequency component on the dc input
current was reduced to 14.6% at 1 kW. It is noted that the remaining
double-line frequency component on the dc input current around thearea of the maximum or minimum output voltage is caused that the
duty of the common mode operation is limited in Fig. 11 (b).
Figure 12 shows the fundamental waveforms with the active
power decoupling method in the lowest input voltage condition at
160 Vdc, which is worst case of the fluctuating input voltage. This
experiment with the low input voltage is operational checks because
it is necessary to take into account the DC input voltage may varydepending on the connected application. Note that the averagevoltage of the buffer capacitor is twice the input voltage, and theother condition in the low input voltage condition at 160 V dc is thesame to the setup as shown in Table 1. From the experimental
results under the steady state, the good active power decoupling
capability is confirmed by the proposed method.

0
vCbuf

Output voltage vout 200V/div.

4 ms/div.

(b) With active power decoupling method.
Fig. 11. Experimental waveforms in steady state when output poweris about 1 kW. The buffer capacitor voltage is absorbing the double-line
frequency power ripple component on the input current.

Buffer capacitor voltage vCbuf 50V/div.
Input current idc 10A/div.
Output current iload 10A/div.
0
0

0
vCbuf

Output voltage vout 250V/div.

4 ms/div.

Fig. 12
Waveforms of proposed isolated converter with active
power decoupling method and low input voltage condition at 1 kW. This
experimental result is limited to compensated capability at the point
between the bottom of the buffer capacitor voltage and the peak voltageof the output side to meet the operation condition in (11).

Figure 13 shows the harmonic analysis result of the input current
ripple component in 200 Vdc input voltage condition of the proposed
converter. The double-line frequency power ripple component as a
100 Hz on the input current is normalized by each DC componentof the input current. As the analysis result, the double-line
frequency ripple component on the input current is reduced by
84.5% compared without the active power decoupling method.
Figure 14 depicts the characteristic of the ripple component
within the output power from 200 W to 2.0 kW. Note that the
7

IEEJ Trans. ●●, Vol.●●, No.●, ●●●96Without active power decoupling method

100

94

Reduced by 84.5%

Efficiency [%]Harmonic component[%](100%: DC current component)

Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)With active power
decoupling method

10
1

90

200 Vdc input voltage
160 Vdc input voltage

88
86
84

0.1

1

2

3 4 5 6 7
Harmonic number

8

9

0

10

300 600 900 1200 1500 1800 2100
Output power [W]

Fig. 15
Measured conversion efficiency of proposed converterwith active power decoupling operation.

Fig. 13
Harmonic analysis result when output power is 1 kW withthe condition as shown in Table 1. The 2nd order harmonic of 100 Hz
component is reduced by 84.5% by proposed converter and the active
power decoupling method.

90

DC-DC con.
conduction loss
DC-DC con.
switching loss
DCDC_Con_loss
Full-brid
Inv.
conduction
loss
DCDC_SW_loss
Full-brid Inv.
FBI_Con_loss
switching
loss
Diode
rectifier
FBI_SW_loss
conduction loss

30
25

Loss [W]

Current ripple(100Hz)[%](100%: DC current component)

92

20

15
10

60

30

Diode_Rectifier

50

0
200 500 800 1100 1400 1700 2000
Output power [W]APD
circuit loss
APD_total_loss
Interleaved con.
Intereaved_Con_loss
conduction loss
Intereaved_SW_loss
Interleaved
con.
switching loss

Proposed
Conventional
1
2
Converter
Converter

Fig. 16
Converter loss analysis with active power decoupling
operation at rated output power of 2 kW. In the proposed converter, theswitching loss of the secondary inverter is almost zero, because it isoperated at the twice grid frequency.

Fig. 14Characteristic of ripple component on input current
within output power from 200 W to 2 kW. The harmonics results of the
proposed converter are confirmed to the good active power decoupling
capability, which is below 75.0% compered by without active power
decoupling over entire power range.

elements and the switching frequency are used as the analysis
conditions, which compare the conventional active power
decoupling topology as shown in Fig. 2. From analysis result, thetotal loss of the proposed converter is 48.1% lower in comparison
with that of the conventional converter in Fig. 2. In the proposed
converter, the arm current rms value of the primary converter due
to the compensated operation of the buffer capacitor is increased or
decreased, so the total conduction loss of the proposed interleaved
converter increases. However, the total conduction loss of the
proposed interleaved converter with the active power decoupling
capability is almost the same the conventional interleaved converterwith APD circuit. It is note that the switching loss of secondary
inverter in the proposed converter is almost zero. The efficiency
improvement of the proposed converter will be achieved by
reducing the conduction loss of the diode rectifier, which ischanging to a synchronous rectifier operation equipped with theswitching devices.

characteristic shows the experimental results with the proposed
active power decoupling method under the condition in Table 1.
This double-line frequency power ripple component (100 Hz) on
the input current was experimental results by applying the active
power decoupling method. From the analysis result, the double-line
frequency power ripple component was reduced by 87.6%
compared without the active power decoupling operation inmaximum. This is attributed that the power pulsation iscompensated by the active power decoupling operation.
Figure 15 depicts the power conversion efficiency of theexperiment setup as shown in Table 1. The configuration of these
efficiency characteristic is result, which includes the active power
decoupling operation. The input voltage is 160 Vdc and 200 Vdc. The
maximum efficiency with an input voltage condition of 200 Vdc
reaches 94.5% at 1410 W in the propose converter. Note that the
maximum efficiency with the low input voltage condition is 95.0%
because this condition has the lower core loss of the coupledinductor and snubber loss than that with the 200-Vdc operation. The
efficiency improvement is from the reduction in applied voltage tothe coupled inductor and the snubber circuit. As an experimental
result, it is confirmed that the proposed converter under the low
voltage condition has a loss, which is 32.2% lower than that underthe 200-Vdc condition.
Figure 16 shows the loss analysis of the semiconductor elements
at the rated power of each converter. The same semiconductor

6.

Conclusion

This paper proposes a novel active power decoupling circuit thatis integrated with an interleaved boost converter for isolated DC tosingle-phase AC conversion applications. The proposed converter
is controlled to the power conversion and the active power
decoupling operation independently by common-mode operation of
the interleaved inverter. Therefore, in the proposed method the
additional switching devices and the magnetic components for the
8

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)Asia), pp.2395-2402 2015(18) J. Itoh, F. Hayashi, “Ripple Current Reduction of a Fuel Cell for a SinglePhase Isolated Converter Using a DC Active Filter With a Center Tap”, IEEE
Trans. Power Electron., Vol. 25, No. 3, pp. 550-556 (2010)(19) N.Takaoka, H.Takahashi, J.Itoh, “Isolated Single-Phase Matrix Converter
Using Center-Tapped Transformer for Power Decoupling Capability”, IEEE
Transactions on Industry Applications, Vol. 54, No. 2, pp. 1523-1531 (2018)(20) Colonel WM. T. Mclyman, "Transformer and Inductor Design Handbook",(21) N.Takaoka, K. Kusaka, H. Higa, J. Itoh, “Novel DC to Single-Phase Isolated
AC Converter Using Coupled Inductor with Power Decoupling Capability”,
2018 20th European Conference on Power Electronics and Applications(EPE'18 ECCE Europe), 2018, pp. P.1-P.10, (2018)(22) M. Hagiwara, H. Akagi, “A Control Method for Regulating the DC Link
Voltage of a Self-Commutated BTB System”, IEEJ Transaction of Industry
Applications, Vol.123, No.1 (2013)active power decoupling operation do not required to employ the
coupled inductor. The performance of the proposed isolated
converter is demonstrated by the experimental result in order toconfirmed the validify of the proposed method. As the experimental
results, the second-order harmonic due to the ripple component of
the dc input current is reduced by 84.5% owing to the active power
decoupling capability. Moreover, the proposed isolated converter
has realized the maximum efficiency of 94.5%.

References
(1)

(2)

(3)

(4)

(5)

(6)

(7)

(8)
(9)
(10)

(11)

(12)

(13)

(14)

(15)

(16)

(17)

Xiaohu Liu, Hui Li, “A new three-phase high-power soft-switched DC-DC
converter based fuel cell power conditioning system with minimized DC
capacitor”, Energy Conversion Congress and Exposition (ECCE) 2013 IEEE,
pp. 4617-4672, (2013)
M. Ehsan Raoufat, Alireza Khayatian, Aslan Mojallal, “Performance
Recovery of Voltage Source Converters With Application to Grid-Connected
Fuel Cell DGs”, Smart Grid IEEE Transactions on, Vol. 9, No. 2, pp. 11971204 (2018)
Junhong Zhang, Jih-Sheng Lai, Rae-Young Kim, Wensong Yu, “High-PowerDensity Design of a Soft-Switching High-Power Bidirectional dc–dc
Converter”, IEEE Transaction on Power Electronics, Vol. 22., No.4 (2007)
Olympio Cipriano da Silva Filho;Bruno Ricardo de Almeida;Demercil deSouza Oliveira Júnior;Tobias Rafael Fernandes Neto, “High-Frequency
Isolated AC–DC–AC Interleaved Converter for Power Quality Applications”,
IEEE Transactions on Industry Applications, Vol. 54, No. 5 (2018)
Shota Kimura, Jun Imaoka, Masayoshi Yamamoto, “Maximum Allowable
Power and Downsizing Performance in High Power Density Boost
Converter”, Journal of the Japan Institute of Power Electronics, Vol. 40, pp.
84-92 (2014)
Yoshiki Kosode, Hidemine Obara, Atsuo Kawamura, Yu Hosoyamada,
Toyoaki Suenaga, Itsuo Yuzurihara, “High Performance Transient Responseof High/Low Pulse Voltage using Two-Phase Interleaved DC-DC Buck
Converter under Half Sampling Time Deadbeat Control”, IEEJ Journal ofIndustry Applications, No.9 Vo.4, pp.444-452 (2020)
A. Ghassemi, P. C. Banerjee, Z. Zhang, A. Hollenkamp, B. Bahrani, “Aging
Effects of Twice Line Frequency Ripple on Lithium Iron Phosphate(LiFePO4) Batteries” 2019 21st European Conference on Power Electronics
and Applications (EPE '19 ECCE Europe), No. CFP19850-ART 2019
T. Shimizu, S. Suzuki: “A single-phase grid-connected inverter with power
decoupling function”, Proc. IPEC 2010, pp. 2918-2923 (2010)
K. H. Chao, P. T. Cheng: "Power Decoupling Methods for single-phase threepoles AC/DC converters", Proc. ECCE2009, pp. 3742-3747 (2009)
Yi Tang, Frede Blaabjerg, Poh Chiang Loh, Chi Jin, Peng Wang, “Decouplingof Fluctuating Power in Single-Phase Systems Trough a Symmetrical HalfBridge Circuit”, IEEE Transaction of power electronics, vol. 30, No. 4 (2015)
Turmandakh Bat-Orgil, Bayasgalan Dugarjav, Toshihisa Shimizu, “Active
Power-Decoupling Circuit to Reduce Ripple Currents of Recycling Batteries
used with Single-Phase Voltage Source Inverters”, IEEJ Journal of Industry
Applications, DOI: https://doi.org/10.1541/ieejjia.L20001041, (2020)
S. Qin, Y. Lei, C. Barth, W. Liu, R. C. N. Pilawa-Podgurski, “A HighEfficiency High Energy Density Buffer Architecture for Power Pulsation
Decoupling in Grid-Interfaced Converters”, IEEE ECCE 2015, pp.149-157(2015)
Chia-Tse Lee, Yen-Ming Chen, Li-Chung Chen, Po-Tai Cheng, “Efficiency
Improvement of a DC/AC Converter with the Power Decoupling Capability”,
IEEE APEC 2012, pp. 1462-1468 (2012)
Jun-ichi Itoh, Tomokazu Sakuraba, Hoai Nam Le, Keisuke Kusaka,“Requirements for circuit components of single-phase inverter applied withpower decoupling capability toward high power density”, 2016 18th
European Conference on Power Electronics and Applications (EPE'16 ECCE
Europe), DOI: 10.1109/EPE.2016.7695444
Keisuke Kusaka, Nagisa Takaoka, Tomokazu Sakuraba, Hiroki Watanabe,
Jun-ichi Itoh, “Single-Phase AC Grid-Tied Inverter with Buck-Type Active
Power Decoupling Circuit Operated in Discontinuous Current Mode”, IEEJ
Journal of Industry Applications, Vol. 10, No. 3, pp. 292-302 (2020)
Ali Tausif, Hoyoung Jung, Sewan Choi, “Single-Stage Isolated Electrolytic
Capacitor-Less EV Onboard Charger With Power Decoupling”, CPSS
Transaction on Power electronics and application, Vol. 4, No. 1, pp. 2019
W. Yao, Y. Tang, X. Zhang, X. Wang, P. C. Loh, F. Blaabjerg, “Powerdecoupling method for single phase differential buck converter” 2015 9thInternational Conference on Power Electronics and ECCE Asia (ICPE-ECCE

Nagisa Takaoka(Student Member) received her B.S. and M.S.
degrees in electrical, electronics and informationengineering from Nagaoka University of Technology,
Niigata, Japan, in 2014 and 2016 respectively. She iscurrently working toward the Ph.D. degree at the
Nagaoka University of Technology, Japan. She is amember of IEEJ and IEEE. Her current researchinterests include matrix converters, dc to singlephase ac converters.

Hiroki Watanabe(Member) received his B.S. and M.S. degrees
from Nagaoka University of Technology, Niigata,Japan in 2013, 2015, respectively. From 2017 to 2018,he was with Swiss Federal Institute of Technology in
Lausanne (EPFL), Switzerland as a trainee. In 2018,
he received his Ph.D. degree in energy andenvironment science from Nagaoka University ofTechnology. From 2018, he has been with Nagaoka
University of Technology as a researcher. His current
research interests include a single-phase power conversion systems andbidirectional isolated DC-DC converters.

Jun-ichi Itoh

(Senior member) received his M.S. and Ph.D.
degree in electrical and electronic systems
engineering from Nagaoka University of Technology,
Niigata, Japan in 1996, 2000, respectively. From 1996
to 2004, he was with Fuji Electric Corporate Researchand Development Ltd., Tokyo, Japan. He was with
Nagaoka University of Technology, Niigata, Japan as
an associate professor. Since 2017, he has been aprofessor. His research interests are matrix converters,
dc/dc converters, power factor correction techniques, energy storage system
and adjustable speed drive systems. He received IEEJ Academic Promotion
Award (IEEJ Technical Development Award) in 2007. In addition, he also
received Isao Takahashi Power Electronics Award in IPEC-Sapporo 2010
from IEEJ, 58th OHM Technology Award from The Foundation forElectrical Science and Engineering, November, 2011, Intelligent Cosmos
9

IEEJ Trans. ●●, Vol.●●, No.●, ●●●Isolated DC-AC Converter with Active Power Decoupling (Nagisa Takaoka et al.)Award from Intelligent Cosmos Foundation for the Promotion of Science,
May, 2012, and Third prize award from Energy Conversion Congress andExposition-Asia, June, 2013. Prizes for Science and Technology(Development Category) from the Commendation for Science andTechnology by the Minister of Education, Culture, Sports, Science andTechnology, April 2017. He is a senior member of the Institute of ElectricalEngineers of Japan, the Society of Automotive Engineers of Japan and the
IEEE.

10

IEEJ Trans. ●●, Vol.●●, No.●, ●●●