Document 200-1

Common Mode Filter Inductor Analysis
Abstract

Noise limits set by regulatory agencies make solutions
to common mode EMI a necessary consideration in the
manufacture and use of electronic equipment. Common
mode filters are generally relied upon to suppress line
conducted common mode interference. When properly
designed, these filters successfully and reliably reduce
common mode noise. However, successful design ofcommon mode filters requires foresight into the nonideal
character of filter components — the inductor in particu­­lar. It is the aim of this paper to provide filter designers
the knowledge required to identify those characteristics
critical to desired filter performance.

Introduction

The filtering of common mode noise is typically not aswell understood as its differential counterpart and thispaper deals with the practical aspects of common mode
filters as related specifically to the common modein­ductor. Common mode noise occurs simultaneously on
both lines of a conductor pair with respect to a common
ground, whereas differential noise occurs between conductor paths. The frequency response characteristics
of filters incorporating different common mode choke
constructions is examined. The filter designer should
then have a better understanding of common mode­inductors and be able to choose the common mode
inductor construction which will yield the required­attenuation characteristic without the additional costof over-designing or the failure of under-designing thecomponent.

ages. An input L-C smoothing filter is generally required
in off-line switching regulators, but these inductors andcapacitors may themselves be sources of EMI. If theinductor is constructed with a relatively high Q material,
it will display substantial ringing and produce spectral
noise energy. Also, switching noise of the converter may
be coupled back into the line through the distributed
capacitance of the inductor. The power transformer may
also ring and couple in ways similar to the filter inductor
and produce its own EMI.
There are semiconductor noise sources associatedwith temperature (thermal noise) within the junction ofdiffer­­​­ing materials (contact noise) and electron-hole
movement in junction devices (shot noise). There exists
low frequency noise attributable to dc current carrying
electronic devices (modulation, flicker, or 1/f noise), due
to the non-ohmic behavior of semiconductors at high fields(hot carrier noise), the generation and recombination of
charge carriers (generation-recombination noise), andinducted noise at the gate of an FET due to the alterationof the source to drain currents by the induced charge at
the gate.2

II. Real vs Observed Filter Frequency Response

Real filters do not follow the theoretical expectations ­provided by standard filter alignments (Bessel,­Chebyschev and Butterworth — See Appendix A). For
example the expected frequency response of a typicalsecond order low pass filter is shown in Figure 1. However, due to the self-resonant behavior of common mode0

I. Types of Noise and Noise Sources

Conducted emissions from power converters are attrib­
utable to a number of causes. The nature of converter­operation (the rectification of the line frequency, andswitching waveforms, for example) and circuit mag­​net­
ics contribute several unique types of noise; also, thecapacitive effects of components and overall mechanicalstructures, such as cases, and the semiconductor­­
components themselves add their own disruptive volt-

10

Attenuation (dB)

Power converters are often major sources of noise inany equipment. Power converters normally produce
common mode and differential mode noise at harmonics of the switching frequency while some wideband ­differential mode noise is usually also produced.1

20
30
40
50
60
70
80
90

0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 1. Ideal frequency response predicted for second
order filter with Coilcraft E3493 common mode ­i­nductor and 0.005 µF capacitors.

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Document 200-2

other windings through a shared core and is undesirable in transformers because it stores energy without­transforming it to other windings in the structure. In a
low pass filter, however, leakage inductance adds tothe attenuation of the filter. In line frequency common
mode chokes (i.e. where the differential signal passes­unattenuated due to coupling of the winds) the uncoupledleakage inductance will aid in the suppression of highfrequency differential noise.

0

Attenuation (dB)

10
20
30
40
50
60
70
80
90

0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 2. Measured frequency response of second
order filter with Coilcraft E3493 common mode inductor
and 0.005 µF capacitors.

inductors (and any inductor), the observed response isactually similar to that shown in Figure 2.
At frequencies below resonance, where the filter components are nearly ideal, standard filter configurations
yield very nearly theoretical results. Inductors, ideally,
provide an impedance which is directly proportional tofrequency, but this is only true for a long, single layer aircore cylindrically wound coil with large gauge wire (i.e.,the ‘perfect’ coil).
At frequencies above self-resonance, an inductor begins
to display the full effects of its parasitic components,­particularly the distributed capacitance (Cd).
The Cd describes the effective capacity across an­­inductor and is caused by individual turns of wire in close
proximity (Figure 3). It is the distributed capacitancethat gives the inductor its characteristic self-resonant
frequency ( 2 1 LC ).
L

R-Series

Resistive losses such as copper (I2R) and core loss­
also affect attenuation. The diameter of wire used in achoke is determined by the amount of current which it­will be required to handle. The larger the current, thelarger the wire. For example, at a line frequency current­of 1 Ampere, 26 AWG wire is required to provide 250­circular mils to support the current. As frequency
increases, the amount of cross sectional area (for asingle strand of wire) used by the current decreases (skin­effect). For frequencies above about 100 kHz, multistranded wire (litz wire, with each strand insulated)should be used if the high frequency current is to besupported. For a low pass inductive filter that needs topass only the line frequencies, further attenuation due
to skin effect is actually desirable.
Capacitors exhibit parasitics of their own. For filter
applications, mylar, mica, and ceramic capacitors are
the most useful because they exhibit high self-resonant
frequencies due to minimization of their parasitics (series
inductance and resistance, and parallel resistance).

III. Winding Configurations

Three winding configurations for inductors are shownin Figure 4. The most simple and least prone to distributed capacitance of all the standard configurations is
the single layer wind. The starts and finishes of a single
layer wind are as far from one another as possible, thus
reducing capacitive coupling. A multilayer wind (twoor more single layers) provides capacitance between
layers as well as from the start lead to the finish lead(that are generally close to one another with the finish
lead ending where the start lead began). The multilayer

1 2 3 4 5 6 7 8 9

9 8 7 6
1 2 3 4 5611
5 4 10 9
1 2 3 7 8 12Single Layer

Double Layer

Progressive

Cd

Figure 3. Model of an inductor (one common mode
inductor winding).

Other non-ideal aspects of inductors include leakage
inductance, which acts as inductance in series with­each winding. All multiple winding chokes display­leak­age inductance. The leakage inductance of a windingis the amount of inductance which is not coupled to any

Figure 4. The three winding configurations examined

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Document 200-3

configuration displays the greatest capacitance of thewinding configurations (thus the lowest attenuation at
higher frequencies for an inductive low pass filter).

In general, above the self-resonant frequency, chokes
become largely dissipative, yielding roughly constantand substantial resistance for a very limited frequency
band (a few octaves at most); above this limited band,the chokes become predominately capacitive (Figure 5).
9080

Attenuation (dB)

70
60
50
40
30
20
10
0
0.10

1.0

Frequency (MHz)

20
0

30
40
50

50 nF
100 nF

60
70
80
90

0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 6. Frequency response of a 2nd order filter with
Coilcraft E3495 for various capacitor values.

IV. Real Data and Description

0.01

Attenuation (dB)

A compromise between the single and multi-layer
winding configurations is the third configuration: theprogressive (banked) wind. A progressive wind isaccomplished by winding a few turns, then backing up a
couple of turns and repeating the process (‘three steps
forward and two steps back’). The start and finish leads
are thus as far apart as possible, with the same numberof turns as a multilayer wind, but without the interleaving.
The distributed capacitance of a progressive winding
configuration is between that of a single layer wind anda multilayer wind.

0
10

10

100

Figure 5. Typical inductor impedance characteristic(Coilcraft E3490).

The self-resonant frequency of a filter’s inductor determines the resonant frequency of the filter itself; maximum
attenuation is normally achieved near this frequency.
The capacitor (in a second order filter configuration)
determines the attenuating behavior of the filter at frequencies above resonance (Figure 6), and with very
little capacitance, the filter response exhibits the rapid
decrease of attenuation due to the inductor response. A
larger value capacitor will increase the slope of attenuation
after resonance for a limited band of frequencies. A verylarge filter capacitor, used to maintain stability out to highfrequencies, will cause the post resonant attenuation toincrease over that achieved at resonance.

Table 1 shows the electrical characteristics of one windof each of the common mode chokes.
Part
E3499
G6252
E3490
E5705
P104
E3493
A-S
A-P
F5806
F5593
B-S
B-P
B-D
F3495
C-D
C-P
C-S
E3506
D-D
D-S
E-S
E-D
E-P

L@10kHz (mH)
20.70
17.70
10.90
6.44
4.00
3.30
2.60
2.50
1.70
1.50
1.20
1.20
1.20
1.08
0.85
0.84
0.80
0.71
0.27
0.27
0.26
0.23
0.21

Freq (MHz)
0.2 to 0.3
0.2 to 0.3
0.3 to 0.4
0.5 to 1.0
0.5 to 1.0
0.5 to 1.0
1.0 to 2.0
1.0 to 2.0
1.0 to 2.0
0.5 to 1.0
1.0 to 2.0
1.0 to 2.0
1.0 to 2.0
1.0 to 2.0
0.5 to 1.0
1.0 to 2.0
1.0 to 2.0
1.0 to 2.0
2.5
4.0
5.0 to 6.0
3.0 to 4.0
5.0 to 6.0

Table 1. Self-resonant frequency ranges and initial
inductance values for the common mode chokes
examined. Note the change in resonant frequency ofthe double layer (-D) compared to the equivalently
constructed single layer (‑S) and progressively
wound (-P) chokes.

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Document 200-4

For similar constructions, self-resonance (the useful andtheoretically predictable frequency limit of inductance)
generally decreases as initial inductance (measured
between 10 kHz and 20 kHz) increases. Moreover, the selfresonant frequency of a layer wound inductor decreases
as the number of layers are increased (while maintaining
the same turn count).

VDif f0.005 µF

LISN

VCM
SOURCELISN

The progressive or banked windings (-P) of the tabledisplay the same self-resonance as the single layer (-S)
versions. Progressive winding allows the increased turnsof a multilayer wind while maintaining the optimum characteristics of a single layer wind.
Because our examination was to determine the attenuation effected by a typical EMI filter configuration due todiffering chokes and choke constructions, we used thecircuit shown in Figure 7, maintaining all circuit components constant except for the inductive element.
0.005 µF

LISN

0.005 µF

RL
OUT

VCM
REF
TEST

VCM
SOURCE

LISN
50 µF

50 µH
50
5

50

5

50
50

50 µH
10 kHz to 2 MHz

50 µH

RL

0.005 µF

50 µH
50
5
5
50
50 µH
10 kHz to 100 MHz

Figure 8. Test circuit used to apply common mode signal
and differential (60 Hz) power simultaneously.

The LISN (see Appendix B) of the “power” circuit was
split at 2 MHz between the standard 50 µHenry/5 Ohm
arrangement (below 2 MHz) and the 50 µFarad/50 Ohm
arrangement (providing 50 Ohms above 2 MHz to thenoise source). Splitting the LISN (and later splicing theattenuation curves at 2 MHz) provided a more accuratecomposite LISN than either arrangement alone would
have with the required power components.
Neglecting measurement error (approximately 4 decibels), the differential input did not appear to affect thecommon mode attenuation of the circuit even at highfrequencies (through 10 MHz).
Figures 9 and 10 show the common mode attenuation
by second order filters with the standard chokes and0.005 µF capacitors and using the LISN load.

2 MHz to 100 MHz
0Figure 7. Test circuit used to measure ­common mode signal attenuation.
Attenuation (dB)

We initially felt it necessary to determine whether
differential power (60 Hz) applied to the common mode
circuit would affect the high frequency common mode
attenuation of the filter. Presumably, the differential
operation of the common mode circuit, with inductors
coupled to each other and their relative polarity suchthat their equal and opposite (differentially produced)
flux lines cancel, no inductive reactance should be
encountered by the differential signal and saturation of
the core should therefore not occur from the differential
signal. To prove this we used the circuit shown in Figure
8 with a load, RL, to provide the rated current throughthe choke.

1020

D-S (272 µH)

30
40F5806 (1700 µH)

5060F5705 (6400 µH)

70
80
90

0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 9. Common mode attenuation of second orderfilters with 0.005 µF capacitors and various
toroid inductors.

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0

0

10

10

20

E3495A(1.2 mH)

30

P104 (4.0 mH)

40
50

Attenuation (dB)

Attenuation (dB)Document 200-5

E3400(11 mH)

60
70
80
90

20

D - Double Layer (7.6 pF)

30
40
50
60
D - Single Layer (5.8 pF)

70
80

0.01

0.10

1.0

Frequency (MHz)

10

90
0.01

100

0.10

1.0

Frequency (MHz)

10

100

Figure 10. Common mode attenuation of second orderfilters with 0.005 µF capacitors and various
E core inductors.

Figure 12. The effect on attenuation due to the differing
distributed capacitance of single and double
layered winds.

The attenuation plots vary greatly below 10 MHz andabove 70 MHz, but provide very similar attenuation
between these frequencies. Above 10 MHz, the capacitor
and its anomalies dominate the filter’s attenuation, andabove 70 MHz the distributed capacitance of the chokeshunts increasingly more noise signal across the choke
as frequency increases. If the distributed capacitanceof a choke is increased (e.g., by multi-layer) the filter
attenuation further decreases and displays this decreaseat lower frequencies.

When comparing the attenuation data of the three
configurations (S=single layer; P=progressive wind;
D=double layer), the difference between the single andprogressive wound chokes are not very significant;the double layer versions do, however, display less­attenuation at high frequencies than either the p­ rogressiveor single layer chokes.

Figures 11 and 12 display the effect of increased distributed capacity (by double layering) of various chokes andconstructions.

0

S (4.8 pF)
P (4.4 pF)

40
50
60
70

10

Attenuation (dB)

Attenuation (dB)20

20

Leakage = 6.6 µH

30
40
50
60
Leakage = 30.6 µH

70
80

80
90
0.01

0

D (16.7 pF)

1030

Figure 13 shows the effect of leakage inductance on
attenuation. It was expected that increased leakage
inductance may slightly increase the high frequency­
common mode attenuation. From the data taken, however,no such trend was apparent, and it appears that typicalchanges in the leakage inductance do not significantly
affect the performance of common mode inductors.

0.10

1.0

Frequency (MHz)

10

100

Figure 11. The effect on attenuation due to thediffering distributed capacitance of single
and double layered winds.

90

0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 13. The effect on attenuation
due to leakage inductance.

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Document 200-6

V. Discussion
The data taken can be used in the following ways:
1. The data show that common mode filter responseis the same as that expected for the more familiar
differential mode L-C filter; common mode filter
response can be fairly accurately predictedusing standard L-C calculations except where thecomponents used exhibit non-ideal characteristics. For example, Figure 2 shows the theoreticallycalculated ­frequency response for an L-C filter
using a 3.3 mH common mode choke, a 0.005 µF
capacitor, and the LISN load. With these component
values one would expect a high frequency rate ofattenuation of 20 dB per decade. It can be seenfrom the actual response of the same filter (shownin Figure 2) that the measured rate of attenuation
for frequencies below resonance (within an octaveof resonance) agrees quite well with the expected
slope and approximate value. This is useful in thatfor frequencies below inductor self-resonance, thecomponent values necessary to achieve a desired
level of attenuation may simply be calculated.
2. The data also show that common mode filters
achieve a maximum value of attenuation at the selfresonant frequency of the common mode inductor.
The self-resonant frequency of the inductor thus
becomes an easily used indicator of whether one
should adjust the capacitor value or the inductor
value to achieve greater attenuation at a specificfrequency or frequency band. For example, Figure
6 shows the differences in attenuation caused bychanging the filter capacitor value. If one wereinterested in attenuation at 4 MHz, it is seen that theattenuation can be increased from 55 dB to 85 dB
by increasing the capacitor value from 50 nF to
100 nF; whereas even a relatively large change in­inductance would have negligible affect.
3. In general common mode filter response can be
broken down into three frequency regions of interest:(A) the region below inductor self-resonance in whichcalculations based on component values hold true;(B) the region near inductor self-resonance in which
the filter achieves the maximum attenuation; and (C)the region above inductor self-resonance in which
the response is dominated by the filter capacitor.

Conclusions
Common mode filter response differs substantiallyfrom theoretically predicted performance. Filter performance can be explained and successfully manipulated

if ­nonideal component response is taken into account.
The common mode inductor is a primary component indetermining the response of a typical filter circuit. The
common mode inductor affects the magnitude (maximum attainable attenuation) and the shape (resonant­frequency) of the frequency response of the filter.
A filter designer should carefully consider filter response
from approximately 1 MHz to 30 MHz to determine
whether the slightly diminished attenuation of a multiple
layer inductor is acceptable. If one were to specify the
inductor to be single layer it may result in unnecessary costand size penalties. The winding pattern of the individual­windings (Cd) is far more significant than the relationshipof the two windings (leakage inductance).
The distributed capacitance of a choke decreases attenuation at high frequencies and multilayering increases
the distributed capacitance of an inductor. Progressive
winding allows the equivalent number of turns of wire asa multilayer and usually far more turns than a single layer
could accommodate. A progressively wound inductor will
display a distributed capacitance similar to a single layer.
To attenuate noise voltages which occur above the limitsprescribed by the FCC and VDE, the filter designer may
opt for the progressively wound inductor.

Appendix A

Passive filters serve as a very good means of eliminating
the majority of conducted noise into a device (or out­­of a device and back into the line) when relatively highcurrent is flowing. Many filter configurations exist andeach has its own advantages. Commonly consideredfilter alignments are the Chebyshev, the Butterworth,and the Bessel.
The ideal Chebyshev low pass filter alignments allow acompromise between the amount of ripple in the passband (and damping) and the slope of attenuation at thecutoff frequency. The stable behavior of the time responseof Chebyschev filters is related to the damping factor, theallowed ripple, and the slope of attenuation at the cutoff
frequency; as the slope of the attenuation at the cutoff isincreased, the transient response becomes less stableand prone to ringing, and the phase response becomesmuch less linear.
Butterworth low pass filter alignments are Chebyshev
filters designed for minimum ripple and ideally provide aflat response, no attenuation prior to the cutoff frequency,
and a damping factor of approximately 0.7. After thecutoff frequency, the attenuation begins and continues
at 20 times the order in decibels per decade. The time
response for the Butterworth filter exhibits some ringing

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Document 200-7

In their use as noise suppressors, filters must be able toeliminate as much noise as possible over a predeterminedfrequency band. As long as the inherent ringing of thefilter is low (does not itself become a noise source) it is
the frequency rather than the time response of a filter
that is an important concern. In a common mode filter,the differential signal does not encounter the filter, thus
any problems associated with phase or time response
affect only the common mode noise. When the filter isused to keep noise from entering the power mains fromthe device, phase and time responses become trivial toeven the differential voltage.
Butterworth alignments for the design of EMI filters arean appropriate starting point; they are easily modeled,readily approximated and constructed, and provide goodfrequency response with relatively little ringing.

90
80
70

Impedance ()and the phase response is not ideal, but predictable. The
overall response of Butterworth filters is well suited toquick and easy methods of approximation. Ideally, Bessel
filters maintain a very stable, linear phase response butat the expense of frequency response; the time responseof Bessel filters is well behaved without substantial overshoot or ringing.

60
50
40
30
20
10
0
0.01

0.10

1.0

Frequency (MHz)

10

100

Figure 14. LISN characteristic specified by FCC Docket
20780 and VDE 0876 Part 1.

(1) Schneider, Len M. “Noise Sources Equivalent Circuit
Model for Off-Line Converters and Its use in Input Filter
Design.” Proceedings of Powercon 10, March 1983, P. C-1.
(2) Scidmore, A.K. “Noise in Amplifiers.” Lecture Notes
for ECE-341—Linear Active Circuits, University of Wisconsin: 1982.
(3) Nave, Mark “Line Impedance Stabilizing Networks:
Theory and Use.” RF Design,April 1985, P. 54.

Appendix B

A major object of testing a component part or device
is the ability to repeat the testing with the same results
regardless of the time or place of the test. When testing
requires the power line to be stable, or at least standard
for a test, adjustments must usually be made or spurious
results may occur.
Power lines may vary in output impedance by as muchas 40 Ohms from location to location, making the­­repeatability of line based evaluations very unreliable
between differing locations. A Line Impedance Stabilizing
Network (LISN, or Artificial Mains Network: AMN) allows
uniform line based testing regardless of locality. A line
impedance standard has been devised and is specified
by several licensing and independent safety agencies(e.g., the FCC and VDE; see Figure 14).3

Kendall, C.M. and Schmid, A.A. “Characteristics and Control of EMI in a Switching Regulator Power Converter.”Proceedings of Powercon 10, March 1983, P. C-4.
Head, Mike “Know RF-Emission Regulations Pertaining
to Your Design.” EDN, August 1981.
Kociecki, John “Predicting the Performance of CommonMode Inductors.” Powerconversion International, March
1984, P. 56.
Carsten, Bruce W. “Design Techniques for the Inherent
Reduction of Power Converter EMI.” Proceedings ofPowercon 11, March 1984, P. D-2.
Nye, J.F. Physical Properties of Crystals. Oxford University
Press, 1972.
Williams, Arthur B. Electronic Filter Design Handbook.
McGraw-Hill, 1981.

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