Q4
2019

NEW PRODUCT
GUIDE
NEW
NEW
NEW

NPG19Q4

TABLE OF CONTENTS

6
14

32

4
16
44

10
34

30

ADAPTERSFILTERSSPLITTERS/COMBINERSAMPLIFIERS
CABLESSWITCHESTRANSFORMERSCOUPLERSDESIGNER'S KITS

AD ADAPTERS
50Ω DC to 40 GHz

NMD 2.92mm and 2.4mm Adapters• Facilitates direct connection to lab instrumentation• Ultra-wideband• Flat response• Low insertion loss
• Excellent VSWR▼

Conn. 1Frequency Range
(MHz)

VSWR
(:1)

KF-24MNMD+

2.92mm-F

2.4mm NMD-M

DC-40000

1.06

KFNMD-24MNMD+

2.92mm NMD-F

2.4mm NMD-M

DC-40000

1.08

KFNMD-KMNMD+

2.92mm NMD-F

2.92mm NMD-M

DC-40000

1.05

KM-24MNMD+

2.92mm-M

2.4mm NMD-M

DC-40000

1.04

KMNMD-24MNMD+

2.92mm NMD-M

2.4mm NMD-M

DC-40000

1.06

50Ω DC to 67000 MHz

1.85mm Adapters• Flat response• Low insertion loss
• Excellent VSWRH I G H L I G H T S▼

Conn. 1

 ew NMD 2.92mm and 2.4mm adaptersNfacilitate connections to lab instrumentation
New 1.85mm adapters up to 67 GHz

Model Number

Conn. 1Conn. 1Frequency Range
(MHz)

VSWR
(:1)

185F-185F+

1.85mm-F

1.85mm-F

DC-67000

1.05

185M-185F+

1.85mm-M

1.85mm-F

DC-67000

1.04

185M-185M+

1.85mm-M

1.85mm-M

DC-67000

1.04

185F-24F+

1.85mm-F

2.4mm-F

DC-500001.08

185F-24M+

1.85mm-F

2.4mm-M

DC-50000

1.08

185M-24F+

1.85mm-M

2.4mm-F

DC-50000

1.06

185M-24M+

1.85mm-M

2.4mm-M

DC-50000

1.04

185F-KF+

1.85mm-F

2.92mm-F

DC-40000

1.05

185F-KM+

1.85mm-F

2.92mm-M

DC-40000

1.04

185M-KF+

1.85mm-M

2.92mm-F

DC-40000

1.04

185M-KM+

1.85mm-M

2.92mm-M

DC-40000

1.03

www.minicircuits.com 5

AMPLIFIERSAMPLIFIERS

S
R
E
T
P
A
D
A

Model Number

S
R
E
I
F
I
L
P
M
A

▼

▼

▼

H I G H L I G H T S

New Ultra-Wideband Coaxial
Amplifers up to 43.5 GHz
New MMIC amplifiers with low noiseand high dynamic range
Wideband class A/AB amplifiers up to 5W

www.minicircuits.com 7

AM AMPLIFIERS50Ω 0.05 to 43500 MHz

50Ω DC to 18000 MHz

Ultra-Wideband Coaxial Amplifiers

MMIC Amplifiers

8• Wideband• Low Noise• High dynamic range

NEW RELEASES
Model Number

Frequency
Range
(MHz)

Gain (dB)
Typ.

NF (dB)Typ.

P1dB (dBm)
Typ.

OIP3 (dBm)
Typ.

Voltage(V)

DC Current(mA)

Model Number

Frequency
Range
(MHz)

Gain
(dB)

NF
(dB)

P1dB
(dBm)

OIP3
(dBm)

Voltage(V)

DC
Current
(mA)

Case
Style

ZVA-24443G1+

24000-43500

45

1.7

20

27

15

160

PMA2-183LN+

4000-18000

10.4

2.5

14.2

25.6

5

48.2

2x2mm QFN

ZVA-02443HP+

2000-43500

37

3.5

17

25

15

130

PHA-83W+

50-8000

15.7

3.3

23.3

35.5

5.0/9.041/110

SOT-89

ZVA-443HGX+

10-43500

33

3.5

9

18

15

225

PSA-39+DC-6000

23

2.2

10.7

23.3

5

32

SOT-363

ZVA-443X+

0.05-43500

11

4.5

10

22

5

80

PSA-8A+

DC-4000

31

3

12.8

25.8

5

36

2x2mm QFN

ZVE-403-K+

26000-40000

22

9

19

21

12

300

ZVA-403GX+

0.05-40000

11

4.5

11

21

5

100

ZVE-323LN-K+

18000-32000

20

3

10

23

12

50

ZVA-203GX+

1500-21000

29

3

15.5

27.5

5

450

ZVA-213-S+

800-21000

26

3

24

33

12

400

ZX60-06203LN+

6000-20000

18.4

2.8

15.6

27.4

5

128

ZX60-24-S+

5000-20000

24

6.8

18

27

5

260

ZX60-24A-S+

5000-20000

24

6.4

18.3

25.4

5

27

ZX60-02203+

2000-20000

21.5

6.5

14.6

28

5

155

ZVA-213UWX+

100-20000

14

3

16

29

+12, -5

84

ZX60-183-S+

6000-18000

23.5

6.9

18.1

27.2

5

ZX60-183A-S+

6000-18000

28

5

18

27

ZX60-06183LN+

6000-18000

25

2.1

11

ZVA-183-S+

700-18000

26

3

ZVA-183G-S+

500-18000

38

ZVA-183W-S+

100-18000

ZX60-153LN-S+

50Ω 0.0025 to 6000 MHz

Wideband Coaxial
Class A/AB Amplifiers• Wideband• High gain, up to 45 dB• Excellent linearity
Model Number

Frequency
Range
(MHz)

Gain
(dB)

NF
(dB)

P1dB
(dBm)

OIP3
(dBm)

Voltage(V)

DC
Current(mA)

Connector
Type

260

ZHL-1W-63-S+

600-6000

35

12

30

35

15

1000

SMA

5

260

ZHL-1W-63X-S+

600-6000

35

12

30

35

15

1000

SMA

24

5

64

ZHL-2W-63-S+

600-6000

42

12

33

38

28

2000

SMA

24

33

12

400

ZHL-5W-63-S+

600-6000

45

12

37

42

28

3500

SMA

3

25

36

15

770

ZHL-6A+

0.0025-500

25

9.5

22

34

24

350

BNC

28

3

26

34.5

15

625

ZHL-6A-N+

0.0025-500

25

9.5

22

34

24

350

N

500-15000

16

2.8

15

27

12

82

ZHL-6A-S+

0.0025-500

25

9.5

22

34

24

350

SMA

ZX60-14012L-S+

0.3-14000

12

5.5

11

20

12

62

ZX60-123LN-S+

500-12000

17

2.4

16

28

12

82

ZX60-05113LN+

5000-11000

21.4

1.9

13

24.5

5

42

ZX60-83LN-S+

500-8000

22.1

1.4

20.7

35.2

5/6

60/70

www.minicircuits.com 9

AMPLIFIERSAMPLIFIERS• Ultra-wideband• Excellent gain flatness• Rugged designs feature built-in protections against
over-voltage, reverse voltage, and in-rush current

S
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L
B
A
C

▼

Flexible interconnect cables▼

H I G H L I G H T S

Ultra-flexible precision test cables

www.minicircuits.com 11

CS CABLES
50Ω DC to 18000 MHz

50Ω DC to 40000 MHz

Flexible Test Cables

Flexible Interconnect Cables• Minimal performance change vs. flexure• Tight bend radius, 2.0”
• Booted joints protect the connector interface during bending• Excellent return loss and insertion loss

• Flexible construction• Ideal for integrating coaxial components and subassemblies• Excellent return loss and insertion loss

12

NEW RELEASES

Length(FT)Conn. 1Conn. 2Item Number

Frequency
Range
(MHz)

Insertion
Loss
(dB)

0.5

ULC-10FT-NMNM+

DC-18000

10.0

N-Type-Male

N-Type-Male

7.0

2.92mm-M

0.6

ULC-10FT-SMSM+

DC-18000

10.0

SMA-Male

SMA-Male

7.5

2.92mm-M

2.92mm-M

0.9

ULC-8FT-SMSM+

DC-18000

8.0

SMA-Male

SMA-Male

5.4

1.0

2.92mm-M

2.92mm-M

1.5

ULC-6FT-NMNM+

DC-18000

6.0

N-Type-Male

N-Type-Male

4.6

0.086

0.5

3.5mm-M

3.5mm-M

0.7

ULC-6FT-SMNM+

DC-18000

6.0

N-Type-Male

SMA-Male

4.4DC-30000

0.086

1.0

3.5mm-M

3.5mm-M

1.29

ULC-6FT-SMSM+

DC-18000

6.0

SMA-Male

SMA-Male

5.2

FL086-6NM+

DC-18000

0.086

0.5

N-Type-Male

N-Type-Male

0.3

ULC-4FT-NMNM+

DC-18000

4.0

N-Type-Male

N-Type-Male

4.0

FL086-6SM+

DC-18000

0.086

0.5

SMA-Male

SMA-Male

0.4

ULC-4FT-SMNM+

DC-18000

4.0

N-Type-Male

SMA-Male

3.1

FL086-6SMNM+

DC-18000

0.086

0.5

N-Type-Male

SMA-Male

0.3

ULC-4FT-SMSM+

DC-18000

4.0

SMA-Male

SMA-Male

3.4

FL086-9SM+

DC-18000

0.086

0.75

SMA-Male

SMA-Male

0.64

ULC-1M-NMNM+

DC-18000

3.28

N-Type-Male

N-Type-Male

2.6

FL086-12NM+

DC-18000

0.086

1.0

N-Type-Male

N-Type-Male

0.6

ULC-1M-SMNM+

DC-18000

3.28

N-Type-Male

SMA-Male

2.5

FL086-12SM+

DC-18000

0.086

1.0

SMA-Male

SMA-Male

0.9

ULC-1M-SMSM+

DC-18000

3.28

SMA-Male

SMA-Male

2.8

FL086-12SMNM+

DC-18000

0.086

1.0

N-Type-Male

SMA-Male

0.7

ULC-3FT-NMNM+

DC-18000

3.0

N-Type-Male

N-Type-Male

2.2

FL086-24NM+

DC-18000

0.086

2.0

N-Type-Male

N-Type-Male

1.4

ULC-3FT-SMNM+

DC-18000

3.0

N-Type-Male

SMA-Male

2.2

FL086-24SM+

DC-18000

0.086

2.0

SMA-Male

SMA-Male

1.5

ULC-3FT-SMSM+

DC-18000

3.0

SMA-Male

SMA-Male

2.7

FL086-24SMNM+

DC-18000

0.086

2.0

N-Type-Male

SMA-Male

1.4

ULC-2FT-NMNM+

DC-18000

2.0

N-Type-Male

N-Type-Male

1.6

FL141-6NM+

DC-18000

0.141

0.5

N-Type-Male

N-Type-Male

0.2

ULC-2FT-SMNM+

DC-18000

2.0

N-Type-Male

SMA-Male

1.5

FL141-6SM+

DC-18000

0.141

0.5

SMA-Male

SMA-Male

0.3

ULC-2FT-SMSM+

DC-18000

2.0

SMA-Male

SMA-Male

1.8

FL141-6SMNM+

DC-18000

0.141

0.5

N-Type-Male

SMA-Male

0.2

ULC-1.5FT-SMSM+

DC-18000

1.5

SMA-Male

SMA-Male

1.4

FL141-9SM+

DC-18000

0.141

0.75

SMA-Male

SMA-Male

0.37

ULC-1FT-SMSM+

DC-18000

1.0

SMA-Male

SMA-Male

0.7

FL141-12NM+

DC-18000

0.141

1.0

N-Type-Male

N-Type-Male

0.4

FL141-12SM+

DC-18000

0.141

1.0

SMA-Male

SMA-Male

0.5FL141-12SMNM+

DC-18000

0.141

1.0

N-Type-Male

SMA-Male

0.4

FL141-24NM+

DC-18000

0.141

2.0

N-Type-Male

N-Type-Male

0.9

FL141-24SM+

DC-18000

0.141

2.0

SMA-Male

SMA-Male

1.0

FL141-24SMNM+

DC-18000

0.141

2.0

N-Type-Male

SMA-Male

0.9

Center
Diameter(IN)

Length(FT)Conn. 1

Item Number

Frequency
Range
(MHz)

FL086-3KM+

DC-40000

0.086

0.25

2.92mm-M

2.92mm-M

FL086-4KM+

DC-40000

0.086

0.33

2.92mm-M

FL086-6KM+

DC-40000

0.086

0.5

FL086-12KM+

DC-40000

0.086

FL086-6-35M+

DC-30000

FL086-12-35M+

Conn. 2Insertion
Loss
(dB)

www.minicircuits.com 13

CABLESCABLESNEW RELEASESCO COUPLERS
50Ω 500 MHz to 50 GHz

Ultra-Wideband Directional Couplers• Ultra-wide bandwidth, now up to 50 GHz• Excellent coupling flatness• Good directivity• DC passing
NEW RELEASES
Model Number▼

H I G H L I G H T S

Ultra-wide bandwidth, now up to 50 GHz

Coupling (dB)Nom.

Mainline
Loss (dB)Directivity
(dB)

VSWR
(:1)

Power Input
Max.
(W)

ZCDC30-K1844+

18000-40000

30

0.6

22

1.15

20

ZCDC30-K0644+

6000-40000

30

0.5

22

1.12

20

ZCDC30-18263-S+

18000-26500

30

0.6

21

1.14

20

ZCDC30-06263-S+

6000-26500

30

0.6

23

1.12

20

ZCDC30-02263-S+

2000-26500

30

0.6

23

1.14

20

ZCDC30-01263-S+

1000-26500

30

0.8

23

1.14

20

ZCDC30-5R263-S+

500-26500

30

0.6

28

1.07

20

ZCDC20-K1844+

18000-40000

20

0.7

19

1.17

20

ZCDC20-K0644+

6000-40000

20

0.7

22

1.07

20

ZCDC20-K0244+

2000-40000

20

1.0

20

1.17

20

ZCDC20-K0144+

1000-40000

20

1.2

20

1.20

20

ZCDC20-18263-S+

18000-26500

20

0.4

24

1.14

20

ZCDC20-06263-S+

6000-26500

20

0.5

26

1.14

20

ZCDC20-02263S+

2000-26500

20

0.5

18

1.33

20

ZCDC20-01263-S+

1000-26500

20

0.9

23

1.12

20

ZCDC20-5R263-S+

500-26500

20

0.9

25

1.09

20

ZCDC16-K1844+

18000-40000

16

0.7

23

1.10

20

ZCDC16-K0244+

2000-40000

16

1.2

20

1.19

20

ZCDC16-K0144+

1000-40000

16

1.3

20

1.22

19

ZCDC16-K5R44W+

500-40000

16

2.0

19

1.19

20

ZCDC16-02263-S+

2000-26500

16

0.6

26

1.09

20

ZCDC16-01263-S+

1000-26500

16

0.9

21

1.14

20

ZCDC16-5R263-S+

500-26500

16

1.4

23

1.12

20

ZCDC13-K26344+

26500-40000

13

0.9

21

1.22

20

ZCDC13-K1844+

18000-40000

13

0.9

21

1.13

20

ZCDC13-K0244+

2000-40000

13

0.95

24

1.11

20

ZCDC13-K0144+

1000-40000

13

1.5

19

1.73

13

ZCDC13-01263-S+

1000-26500

13

1.2

21

1.17

19

ZCDC13-5R263-S+

500-26500

13

1.3

21

1.73

20

ZCDC10-V254+

2000-50000

10

1.3

23

1.08

13

ZCDC10-K1844+

18000-40000

10

1.2

21

1.22

17

ZCDC10-K0644+

6000-40000

10

1.0

24

1.12

17

ZCDC10-K0244+

2000-40000

10

1.2

23

1.11

15

ZCDC10-K0144+

1000-40000

10

2.2

16

1.22

19

ZCDC10-K5R44W+

500-40000

10

1.3

23

1.12

15

ZCDC10-18263-S+

18000-26500

10

0.9

24

1.15

20

ZCDC10-06263-S+

6000-26500

10

1.0

22

1.17

20

ZCDC10-02263S+

2000-26500

10

0.9

27

1.11

20

ZCDC10-01263-S+

1000-26500

10

0.9

21

1.17

20

ZCDC10-5R263-S+

500-26500

10

1.2

22

1.12

20

www.minicircuits.com 15

COUPLES

S
R
E
L
P
U
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C

Frequency
Range
(MHz)

FL FILTERS
50Ω 2400 to 2500 MHz

LTCC Balun-Filters
Model
NumberPassband(MHz)

Stopband F3(MHz)

Rejection @ F3(dB)

Stopband F4(MHz)

Rejection @ F4(dB)

BFGE2-552R+

4900-5875

DC-3500

49--

BFGE1-252R+

2400-2500

1000-2000

39

4800-5000

49

BFNL2-252R+

2400-2500

1710-1910

38

4800-5000

42

Passband(MHz)

Stopband F3(MHz)

Rejection @ F3(dB)

Stopband F4(MHz)

Rejection @ F4(dB)

ZVBP-11G3-S+

11200-11400

DC-11030

35

11580-20000

35

ZVBP-10R5G-S+

9750-11250

DC-5950

35

15100-18000

35

ZVBP-9R6G-S+

9550-9650

DC-930030

9900-20400

33

ZVBP-8250-S+

8025-8475

DC-765020

8925-11000

20

ZVBP-7100-S+

7025-7175

DC-6990367224-14000

35

ZVBP-5800-S+

5725-5875

DC-520035

6400-14000

35

ZVBP-5310-S+

5250-5370

DC-5080205530-8250

20

ZVBP-4900-S+

4840-4960

DC-467020

5100-9000

20

ZVBP-4810-S+

4750-4870

DC-460020

5020-8250

20

ZVBP-4300-S+

4250-4350

DC-414020

4480-8000

20ZVBP-4000-S+

3997-4003

DC - 3800

70

4200 - 6000

70

LTCC Balun Filters

ZVBP-3875-S+

3845-3905

DC-378535

3970-8500

35

2400-2500

2120-2260

40

2635-2780

40▼

New Cavity Filters

ZVBP-2450-S+
ZVBP-2400-S+

2375-2425

DC-225035

2550-6000

35▼• Integrated balun and bandpass filter in a single 0805 monolith• Band-optimized for Wi-Fi, Bluetooth and Zigbee

Application Note: Filtering without Reflections

ZVBP-2300A-S+

2200-2400

DC-200030

2550-8050

30

ZVBP-2072R5-S+

2030-2115

DC-193048

2220-6000

49

ZVBP-1420-N+

1415-1425

DC-137081

1470-3000

79

ZVBP-909-S+

902-915

10-89520925-2300

20ZVBP-909A-S+

902-915

DC-89527923-2300

3550Ω 902 to 11400 MHz

S
R
E
T
L
I
F

Cavity Bandpass Filters• Passbands as narrow as 1%• Very low passband insertion loss
• Very fast roll-off with wide stopband• High power handling, up to 15W
NEW RELEASES▼H I G H L I G H T S

www.minicircuits.com 17

FILTERS

Model
Number

FILTERING WITHOUTREFLECTIONS:
Flattening Multiplier Chain Conversion
Efficiency & More

Rohan Shrotriya, Mini-Circuits Applications
Dr. Matthew A. Morgan, National Radio Astronomy Observatory

ABSTRACT18

FILTERSA new class of filter, which exhibits
broadband matched impedance at its
ports, has recently been invented andmade available. This new device, thereflectionless filter, has demonstrated avariety of benefits when used to replace
conventional filters in a signal chain.
This white paper briefly introduces
the reflectionless filter and compares
conventional filter and reflectionlessfilter behavior. Use cases are presented
examining how reflectionless filters canimprove system performance when usedwith mixers, ADCs, and receiver signal
chains. Lastly, an experiment is described
and test results presented to compare
the conversion loss ripple in multiplierchains when reflectionless and reflective
filters are used to filter spurious signals.

www.minicircuits.com 19

FILTERING WITHOUT REFLECTIONS:INTRODUCTION

Overview of Reflectionless Filter Theory

The advent of broad bandwidth amplifiers, analog-to-digital converters (ADCs), digital-to-analog converters (DACs),and software-defined radios has brought about growing interest in broadband communications, radar, and sensing
applications. For these applications, there is often a need to preserve the highest sensitivity and dynamic range possible
within the receiver signal chain and to mitigate the number and strength of harmonics and spurious content in thetransmitter signal chain. This is a substantial challenge considering the nature of non-linear components within thesecircuits, namely compression-mode amplifiers, mixers, multipliers, and frequency-conversion electronics.

Reflectionless filter topologies are often realized as symmetric networks. This enables the use of even-/odd-mode analysis
techniques, which aids in understanding the reflectionless nature of these filters. Given a two-port network with a perfectsymmetry plane dividing the circuit, there are two modes useful for analysis. The even mode occurs when both ports arestimulated by signals of equal amplitude and phase, which means there is no current passing across the symmetry plane.
The odd mode occurs when the signals stimulated at either port are equal in amplitude and opposite in phase (180degrees out of phase), and where the voltage potential at the nodes along the symmetry plane is 0 referenced to ground.

A long-overlooked opportunity to enhance the signal-to-noise ratio (SNR) and dynamic range within a signal chain
and to reduce harmonics/spurious content within these circuits is to address a seemingly innate property of filters: theirout-of-band reflective behavior. Reflectionless filters utilize a novel circuit topology to effectively eliminate the standing
waves created by traditional filters without additional components (such as pads). This unique property gives designers
a new way to improve the system performance of a wide array of broadband circuits, or any circuits suffering fromout-of-band impedance mismatch.

Under these conditions, two separate single-port networks can be drawn, each with only half the elements of the original
two-port network with the nodes along the symmetry plane either shorted to ground or open (odd-mode circuits andeven-mode circuits). Hence, the scattering parameters of the two-port network can be derived from the superposition
Equation 1:of the reflection coefficients of the two circuits detailed in Equations 1 and 2.

REFLECTIONLESS FILTER BASICS

A Brief History of the Genesis of Reflectionless Filters

In the case of non-linear devices (amplifiers, mixers, converters, etc.), this standing wave can lead to a number ofundesirable effects including:• Gain compression• Oscillations• Proliferation of spurious mixing products• Unexpected resonances• Degraded stability

Γ"#"$ = 𝑠𝑠'' + 𝑠𝑠') = 𝑠𝑠)) + 𝑠𝑠)' & Γ,-- = 𝑠𝑠'' − 𝑠𝑠') = 𝑠𝑠)) − 𝑠𝑠)'
Equation 1:
Equation 2:
Γ"#"$ = 𝑠𝑠'' + 𝑠𝑠') = 𝑠𝑠)) +Equation𝑠𝑠)' & 2Γ,-- = 𝑠𝑠'' − 𝑠𝑠') = 𝑠𝑠)) − 𝑠𝑠)'
Equation 2:
1
1

𝑠𝑠'' = 𝑠𝑠') = (Γ"#"$ + Γ,-- ) & 𝑠𝑠)' = 𝑠𝑠') = (Γ"#"$ − Γ,-- )12
12𝑠𝑠'' = 𝑠𝑠') = (Γ"#"$ + Γ,-- ) & 𝑠𝑠)' = 𝑠𝑠') = (Γ"#"$ − Γ,-- )Equation 2:2
2
Given the S-parameters, a condition of1
exact input match (S11 = S22 = 0) can be derived
1 from Equation 3. This condition
Equation
3:𝑠𝑠if the=even-mode
can only besatisfiedandodd-mode
circuitsareduals
of eachother(i.e. where
their)reflection
(
)
𝑠𝑠
=
Γ
+
Γ
&
𝑠𝑠
=
𝑠𝑠
=
(Γ"#"$− Γ,-''
')
"#"$
,-)'
')
coefficients are additive inverses of one2another), which means that the inductors andcapacitors, along with the series
2Equation 3:
𝐹𝐹𝐹𝐹𝐹𝐹circuit𝑠𝑠'' are= swapped𝑠𝑠)) = 0,𝑠𝑠)'= 𝑠𝑠other.
Γ"#"$Γ,-and shunt elements within eachcomparedto theThe
result =of thecombined circuit is a') =transfer function for the original two-port network equivalent to the reflection coefficient of the even-mode circuit.
𝐹𝐹𝐹𝐹𝐹𝐹 𝑠𝑠'' = 𝑠𝑠)) = 0,𝑠𝑠)' = 𝑠𝑠') = Γ"#"$ = Γ,-Equation 3:

Equation 3

𝐹𝐹𝐹𝐹𝐹𝐹 𝑠𝑠'' = 𝑠𝑠)) = 0,𝑠𝑠)' = 𝑠𝑠') = Γ"#"$ = Γ,--Therefore, low-pass, high-pass, band-pass, and band-stop filters can be generated by choosing a filter topology whosereflection characteristic matches the desired transfer characteristic (even-mode circuit), creating a dual of the chosen filter
topology (odd-mode circuit), performing topological modifications that result in symmetry between the even-/odd-mode
circuits without affecting the circuit behavior, then combining the two circuit-halves to form the final two-port filter
network (See Figure 1 and 2).

• Degraded dynamic range• Greater susceptibility to process tolerances/environmental factors• Biasing issues
Hence, it is often necessary to insert isolators or attenuators (pads) into the signal path to dampen this standing
wave effect. However, neither isolators nor pads are an ideal solution for most applications. Isolators are band-limited
devices with a large footprint, while attenuators are broadband absorbers that absorb even pass-band signal energy.
The increased insertion loss from either of these methods may lead to the need for additional gain in the system,
and both solutions add cost, size, complexity, and failure points to the circuit.
A better solution is to design a filter that absorbs signal energy in the stop-band instead of reflecting it. Though absorptive
filter technologies have existed for some time, their design and implementation didn’t meet all of the desired criteria for
a true conventional filter replacement. Hence, a ground-up design effort culminated in the birth of the reflectionless filter,
which is a set of filter topologies and designs that inherently exhibit a broadband matched impedance.

20

Figure 1: (a) A ladder topology high-pass filter is converted to an even-mode equivalent circuit for a low-pass reflectionless filter.
(b) The inductors in red are eliminated in the open circuit case and the capacitors in blue are eliminated during a short circuit casewhen the plane of symmetry is opened/shorted to develop the odd-mode equivalent circuit from the even mode equivalent circuit.
(c) The odd-mode equivalent circuit is developed form the short-circuit case and simplified.

www.minicircuits.com 21

FILTERSFILTERS

Reflectionless filters were born from a desire to enhance the signal chain performance of sensitive radio astronomy
receivers. Innovators of the technology noted that the typical performance of conventional filters (reflective filters) only
exhibited a matched impedance at its ports within the filter’s pass-band. The stop-band regions of these filters areintentionally designed to have very poor impedance match. As a result, undesired stop-band signals, including harmonics,
interference, and noise, are all reflected from the filter ports back through the signal chain. If these unwanted signals arereflected back to another reflective device, a standing wave effect emerges. This standing wave will persist and build on
itself until the attenuation of the transmission path between the two reflective components dampens and absorbs theunwanted signal energy.

Equation 1:
Γ"#"$ = 𝑠𝑠'' + 𝑠𝑠') = 𝑠𝑠)) + Equation𝑠𝑠)' & 1Γ,-- = 𝑠𝑠'' − 𝑠𝑠') = 𝑠𝑠)) − 𝑠𝑠)'FILTERING WITHOUT REFLECTIONS:

Figure 4: A compact Reflectionless Filter package built using a gallium arsenide (GaAs) passive device process.

As with any practical filter, additional loss and parasitics do affect performance relative to the ideal filter circuit. In thecase of the GaAs IPD process, the extremely small filters are built with equally compact and planar spiral inductors
and capacitors, which have relatively low Q values. Hence, the tradeoff for extremely compact size (3x3 QFN package)and excellent repeatability in these filters is generally greater characteristic insertion loss. However, a benefit of thereflectionless filter topology is that these filters are readily cascadable, so sharper roll-off and greater stop-band
rejection may be achieved by adding filters as modular building blocks.
This cascadability allows designers to combine reflectionless filters to achieve a desired response. For example,
reflectionless filters may be cascaded to enhance a filter’s stop-band attenuation, extend the stop-band match, or create
ultra-wideband (UWB) filters. The examples plotted in Figure 5 show that the stop-band attenuation, which is ultimately
limited by the fabrication constraints of the GaAs IPD process, can be improved by combining two complementary
low-pass reflectionless filters. Moreover, Figure 6 illustrates that a UWB band-pass filter with good stop-band match
can be achieved by combining a low-pass and a high-pass reflectionless filter.
S-Parameters of a Commercially Available Low-passReflectionless Filter (XLF-112H+)
S11

0

-20

-20
Sij (dB)

Sij (dB)

It is important to note that for some filter responses, such as Chebyshev, the filter ripple is bounded by the need tomaintain positive element values (unless a modification is applied which requires the use of transformers [6]). Hence,for these types of filters there is an additional limit to the stop-band rejection, which means the stop-band rejection
only reaches approximately 13-14 dB and is somewhat dependant on the order of the filter. Therefore, when using
these filter types to design reflectionless filters it may be necessary to cascade several filters or use a filter response
less encumbered by this restriction (i.e. Zolotarev response).

S11

S21

0

-40

-60

S22

S21

-40

-60

-80
0.01

0.1

1

-80
0.01

10

0.1

Frequency (GHz)1

10Frequency (GHz)

Figure 5: S-Parameters of a single reflectionless low-pass filter versus two cascaded reflectionless low-pass filters

Figure 3: An example 5th order Chebyshev type 1 reflectionless filter.
S-Parameters of an Ultrawide Bandpass Filter Using Two
Reflectionless Filers (XHF-292M+ & XLF-73+)

S-Parameters of a Commercially Available High-passReflectionless Filter (XHF-292M+)

0

-20

-20
Sij (dB)

Sij (dB)

0

-40

-60

-40

0.5

1

5

10

-60

0.5

Frequency (GHz)

1

5

10

Frequency (GHz)

Figure 6: A UWB band-pass filter with good stop-band match can be realized by combining a low-pass and a high-pass reflectionless filter.

22

A Note on Practical Reflectionless Filters

REFLECTIONLESS FILTERS VERSUS CONVENTIONAL FILTERS

In their current incarnation, reflectionless filters have been mass-produced using a gallium arsenide (GaAs) semiconductor
Integrated Passive Device (IPD) fabrication process (see Figure 4). Reflectionless filters with frequency response fromhundreds of megahertz to over 30 gigahertz have been realized [9, 10}. A benefit of the GaAs IPD process is thatthe resulting reflectionless filters are also extremely stable over temperature and exhibit a relatively high operating
temperature beyond 100 degrees C.

It is easy enough to understand the value of having pass-band and stop-band matched impedance at each port of afilter in concept, but it is more instructive to see practical results of the difference between reflective and reflectionlessfilters in specific implementations. The following examples and discussion revolve around the effects of cascading bothreflective and reflectionless filters. Here we examine the negative performance impacts of poor impedance match,compared to filters that are matched at every frequency.

www.minicircuits.com 23

FILTERSFILTERS

Figure 2: An Nth order high-pass reflectionless filter circuit based on even-odd mode synthesis.

S22

S-Parameters of Cascaded Commercially Available Low-pass
Filters (XLF-112H+ & XLF-332+)

FILTERING WITHOUT REFLECTIONS:

Single XLF-421+ GD

Insertion Loss Comparison of Cascaded Filters (Reflectionlessvs. Reflective)
Single XLF-421+ IL

2 XLF-421+ IL

3 XLF-421+ IL

Insertion Loss Comparison of Cascaded Filters (Reflectionlessvs. Reflective)
Single XLF-421+ IL

Single NLP-550+ IL

2 XLF-421+ IL

Insertion Loss (dB)

Insertion Loss (dB)

100
75
50
25

1000

2000

3000

4000

2 NLP-550+ GD

40

2

20

0
-2
-4

1000

2000

3000

Frequency (MHz)

4000

5000

0

-20

-40

1000

2000

3000

4000

5000

Frequency (MHz)REFLECTIONLESS FILTER USE CASES
The following section discusses several applications where replacing conventional filters with reflectionless filters canimprove overall system response. Finally, a detailed experiment is presented to illustrate the use of reflectionless filters in
multiplier chains and related benefits for system performance.

4

2

0

5000

Pairing with Mixers (Up-/Down-converters) To Improve SNR and Dynamic Range
100

200

300

Frequency (MHz)

400

500

600

Frequency (MHz)corner response due to their poor match in the transition. For example, Figure 8 compares the input and output returnloss of cascaded reflective and reflectionless filters. The input and output return loss of just two conventional filter sections
in cascade varies significantly around the cutoff frequency. By contrast, the three-section cascaded reflectionless filter
exhibits relatively consistent input and output return loss in the pass-band to stop-band transition. This example indicatesthat where cascading conventional filters degrades the filter’s frequency response (i.e. passband performance), cascadingreflectionless filters simply adds a predictable amount of insertion loss without other unwanted effects.
Lastly, the reactive impedance of a conventional filter tends to result in distortion and phase instability, which worsens asfilter sections are added. These performance degradations directly impact the group delay performance of conventionalcascaded filters (see Figure 9) [10]. Cascaded reflectionless filters, on the other hand, exhibit relatively flat group delay
through the pass-band, transition, and stop-band.

One inherent aspect of a mixer’s non-linear behavior is the creation of undesirable leakage and spurious signals in various
areas of the spectrum (images and etc.). Filters are often used to help suppress these unwanted mixer products (seeFigure 10). However, at the ports of a mixer, a conventional filter’s stop-band reactive impedance can degrade dynamic
range and conversion loss. It can also lead to the development of nulls and increase sensitivity to process variations, phaseinstability, amplitude instability, expansion of the intermodulation products, and more. Replacing a conventional filter
with a reflectionless filter in an anti-aliasing or image rejection application can improve sideband separation, enhancecalibration longevity, and mitigate mixer sensitivities to process tolerances. Moreover, a reflectionless filter and mixer pair
(if the reflectionless filter is well chosen for the specific mixer) exhibits nearly the same IP3 performance as the mixer alone,which is not the case for conventional filters. Therefore, replacing a conventional filter with a reflectionless filter may allow
the use of a mixer with lower intrinsic IP3 while still meeting system specifications, or at least avoid the need for a mixer
with higher IP3 to meet the same system specifications.

Figure 8: Comparison of output return loss of cascaded reflective and reflectionless filters.
Figure 9: Comparison of group delay of cascaded reflectionless and reflective filters.

Input & Output Return Loss of Cascaded Filters (Reflective)

Input & Output Return Loss of Cascaded Filters (Reflectionless)
Single XLF-421+ IRL

2 XLF-421+ IRL

3 XLF-421+ IRL

2 XLF-421+ ORL

3 XLF-421+ ORL

Single NLP-550+ IRL

Single XLF-421+ ORL

2 NLP-550+ IRL

Single NLP-550+ ORL

2 NLP-550+ ORL

20

50

15
Return Loss (dB)

Return Loss (dB)

40
30
20
10
0
400

500

600
Frequency (MHz)

24

700

800

10

5

0
400

500

600
Frequency (MHz)

700

800

Figure 10: Using conventional (reflective) filters generates reflections and standing waves from out-of-band signals at each port of the conventional
filters. Signal path for reflectionless filters shown in blue. Signal reflections created by conventional filters shown in red and yellow.

www.minicircuits.com 25

FILTERSFILTERS

Single NLP-550+ IL

6

125

0

3 XLF-421+ IL

2 NLP-550+ IL

2 NLP-550+ IL

Single NLP-550+ GD

3 XLF-421+ GD

4-6

When cascaded, conventional reflective filters also suffer degradation of input and output return loss of the transition

2 XLF-421+ GD

Group Delay (ns)Figure 7: Cascading conventional filters increases ripple in the stop-band and passband. Reflectionless filters eliminated this unwanted
effect but introduce an expected accumulation of insertion loss.

Group Delay of Cascaded Filters (Reflective)

Group Delay of Cascaded Filter (Reflectionless)Group Delay (ns)

Whereas cascading conventional filters can lead to ripple and cause phase instability in the pass-band, these same
concerns are not applicable to reflectionless filters. For example, when comparing conventional filters with reflectionlessfilters in series, the cascaded conventional filter response will likely suffer from increased ripple in the stop-band due toan unstable phase relationship between the through and reflected signal. Moreover, a conventional filter will also sufferfrom additional pass-band ripple as a product of the resulting reflections in the transition. Cascading reflectionless filters,by contrast, eliminates the distortion and additional ripples common to conventional filters, but creates an expected
accumulation of insertion loss (see Figure 7) [10].

FILTERING WITHOUT REFLECTIONS:
Wideband ADC Antialiasing Filter
During the sample-and-hold or track-and-hold operations of an ADC, the switching action of the input stage creates a rapidchange in the load conditions for the driving circuit. These switching transients may also generate pulses that back-propagate
through the system. Though intended for this purpose, the switching transients and other non-linear products may occur
at frequencies well beyond the anti-aliasing filter’s operating frequency range. As these filters aren’t intended to be true
broadband devices, the impedance matching at these higher frequencies may be poor enough to yield reflections of theswitching transients and non-linear products back to the ADC input, create a standing wave and otherwise impact the
ADC’s output performance (spur-free dynamic range and noise figure).
In this case, reflectionless filters can be employed at the input of the ADC to absorb the out-of-band signals and mitigate theeffect of switching transients. Reflectionless filters can be added to both differential inputs of the ADC, eliminating differential andcommon-mode switching transients (see Figure 11). If greater selectivity is required than what the typical MMIC reflectionlessfilters can offer, a traditional anti-aliasing filter with higher Q elements or a cascaded reflectionless filter could be added in series.

Multipliers and Multiplier Chains: Stabilizing Conversion Efficiency over Frequency

EXPERIMENT
Frequency translation devices such as multipliers and dividers are used to convert frequencies from lower spectrum regimes
to higher frequencies, and vice versa. As these devices are intrinsically non-linear, they generate spurious harmonics, which
are often filtered to prevent harmonics from appearing in-band. Using conventional, reflective filters creates an undesirable
scenario where the out-of-band harmonics are reflected back to the multiplier. The multiplier is also affected by the reactive
loading exhibited by a reflective filter at harmonic frequencies (see Figure 13). Given that multipliers have poor output
return loss, this combination of effects leads to large ripples in the conversion efficiency of a multiplier chain, and hence,
susceptibility to environmental factors.
As discussed earlier, this issue can be solved by leveraging the unique capabilities of reflectionless filters. In order todemonstrate this solution, an experiment was conducted with a doubler test circuit and a 4X multiplier chain. Each experiment
was conducted using comparable reflective and reflectionless filters, and the results were then analyzed (see Figure 14).
The tests were conducted using an Agilent E8257D PSG signal generator and an Agilent U2000A power meter. The
doubler used in all tests was a Mini-Circuits ZX90-2-36-S+. The second doubler was a KSX2-24+. The amplifier used in
the multiplier chain was a GALI-39+. The conventional filter was a Mini-Circuits VLF-6400+, and the reflectionless filter wasa Mini-Circuits XLF-662M+. Each test was done using four different lengths of interconnect between the connectorized
doubler and filter: no cable, a 6-inch coaxial cable, a-12 inch coaxial cable, and a 36-inch coaxial cable. An image of thetest setup of the doubler-filter pair is presented in Figure 15.

Optimizing Receiver Sensitivity and Dynamic Range with a Distributed Filter Approach
Optimizing a receiver signal chain is essential to achieving the desired sensitivity and dynamic range for the system.
As is often the case with RF circuits, there is generally a trade-off between sensitivity and dynamic range when attempting
to filter out-of-band signals.
Filtering prior to amplifying the received signal attenuates the out-of-band noise and interference, improving dynamic
range. However, this approach also attenuates the strength of the received signal, which reduces the overall receiversensitivity. Placing the filter after the amplifier allows for greater sensitivity, but reduces dynamic range. It may therefore
be desirable to distribute filters with a minimal amount of insertion loss throughout the signal chain.
Unfortunately, cascading conventional (reflective) filters isn’t viable as the out-of-band interactions between the filters
can further degrade signal quality. Reflectionless filters, however, can be cascaded without any undesirable side-effects.
Hence, a new modular approach of cascading reflectionless filters throughout a signal chain to reach the desired levelof sensitivity and dynamic range becomes possible (see Figure 12).

Figure 12: Block diagram representing the process of distributing filters throughout a signal chain to optimize response.

26

Figure 13: The reflections that develop into standing waves in a multiplier chain occur at the output of the multipliers and the
input of the conventional reflective filters.

Figure 14: (a) A diagram of the experimental multiplier chain block diagram. (b) A diagram of the experimental signal chain.

Figure 15: (top) A reflectionless filter and doubler pair. (bottom) A reflective filter and doubler pair.

www.minicircuits.com 27

FILTERSFILTERS

Figure 11: Block diagram depicting how a dual-matched reflectionless filter can be used to improve ADC anti-alias filtering.

FILTERING WITHOUT REFLECTIONS:RESULTSXLF-662M+ 0" @4dBm

XLF-662M+ 0" @6dBm

XLF-662M+ 0" @8dBm

VLF-6400+ 0" @4dBm

VLF-6400+ 0" @6dBm

VLF-6400+ 0" @8dBm

XLF-662M+ 0" @10dBm

XLF-662M+ 0" @13dBm

XLF-662M+ 0" @16dBm

VLF-6400+ 0" @10dBm

VLF-6400+ 0" @13dBm

VLF-6400+ 0" @16dBm

25

25

23

23

21
19
17
15
13
11
9
2900

3000

3100

3200

3300

3400

19
17
15
13
11
3000

3100

Frequency (MHz)

3200

3300

3400

3500

Frequency (MHz)Figure 16: Conversion loss of a doubler-filter pair at various power levels. Reflectionless filter (left); Reflective filter (right).
Conversion Loss of Doubler and Reflective Filter Pair At
Several Power Levels and 6" Cable Length

XLF-662M+ 6" @4dBm

XLF-662M+ 6" @6dBm

XLF-662M+ 6" @8dBm

VLF-6400+ 6" @4dBm

VLF-6400+ 6" @6dBm

VLF-6400+ 6" @8dBm

XLF-662M+ 6" @10dBm

XLF-662M+ 6" @13dBm

XLF-662M+ 6" @16dBm

VLF-6400+ 6" @10dBm

VLF-6400+ 6" @13dBm

VLF-6400+ 6" @16dBm

25

25

23

23

Conversion Loss (dBm)

Conversion Loss (dBm)

FILTERS

Conversion Loss of Doubler and Reflectionless Filter Pair At
Several Power Levels and 6" Cable Length

21
19
17
15
13
11
9
2900

3000

3100

3200

3300

3400

19
17
15
13
11
3000

3100

Frequency (MHz)

3200

3300

3400

3500

Frequency (MHz)Figure 17: Conversion loss of doubler-filter pair with 6-inch cable length between the filter and doubler.
Note the increased transmission length leads to greater development of standing waves.
Conversion Loss of Doubler and Reflectionless Filter Pair At
Several Cable Lengths
XLF-662M+ 0" @8dBm

XLF-662M+ 6" @8dBm

XLF-662M+ 12" @8dBm

VLF-6400+ 0" @8dBm

Conversion Loss (dBm)

Conversion Loss (dBm)

14

12

3000

3100

3200
Frequency (MHz)

3300

3400

3500

Conversion Loss (dB)

15

13
2900

3000

3100

3200

3300

3400

3500

20

REFLECTIONLESS FILTERSKEY FEATURES

18
16
14
12
10
2900

3000

3100

3200

3300

3400

Frequency (MHz)

Lastly, in Figure 19 (right) an X4 multiplier chain is tested using two doublers with a filter after
the first doubler. This plot shows that the conversion loss ripple under worst-case conditions(extended transmission line) is greater for the reflective filter than the reflectionless filter byover 3 dB peak-to-peak. In order to achieve comparable conversion loss ripple with aconventional filter to that of a reflectionless filter, a much higher power amplifier and an extra
attenuator would be necessary to push the doublers well into saturation and mitigate thestanding wave generated by the higher power level. In longer multiplier chains, the reflective
nature of conventional filters may degrade conversion loss flatness even further and require
careful selection of amplifiers and attenuators distributed throughout the multiplier chain,
adding size, cost and complexity to the system. By contrast, a well-chosen amplifier with areflectionless filter can achieve flat conversion loss efficiency while minimizing cost, size andthe need for extra components.

1. Ideal for broadband systems
sensitive to out-of-bandstanding waves
2. Enable integration of
wideband amplifiers
without creating
out-of-band instabilities
3. May be cascaded tocreate UWB filters
4. Ideal for cases where
stop-band signals present
significant harmonics
or interference affecting
the source
5. Reduce BOM byreplacing filter/attenuatorpairs (absorbers)

6. Ideal where stable performanceThis white paper has reviewed the basic genesis and theory of reflectionless filters, comparedover temperature is desirable
conventional reflective filters and reflectionless filter technology (both theoretically andpractically), and described several applications that may benefit from replacing conventional
7. High operating temperaturefilters with reflectionless filters. An experimental example was also described in detail and test
allows for placement close toresults analyzed to illustrate the benefits of using reflectionless filters with multipliers and in
high-power components
multiplier chains. This discussion has shown several ways that reflectionless filters can overcome
many of the typical design challenges associated with introducing reflective filters into a signal
chain, and in particular the development of standing waves from stop-band signals due tothe reactive loading of reflective filters. Reflectionless filters have become a valuable new tool in the RF/microwave engineer’s
toolbox, and future developments of reflectionless filter technology may yield even higher performing filtering solutions.

2. M. Morgan, Reflectionless Filters, Norwood, MA: Artech House, January 2017.
3. Hong J, (ed.). Advances in Planar Filters Design. Institution of Engineering and Technology, 2019. 424 p.

16

4. M organ, M. A., & Boyd, T. A. (2011). Theoretical and Experimental Study of a New Class of Reflectionless Filter.
IEEE Transactions on Microwave Theory and Techniques, 59(5), 1214–1221. doi:10.1109/tmtt.2011.2113189

14

12

10
2900

17

22

1. M. Morgan, “A better way to filter -- part I,” NRAO Blog, March 29, 2018.

VLF-6400+ 12" @8dBm

18

16

10
2900

VLF-6400+ 6" @8dBm

24
19

REFERENCES

Conversion Loss of Doubler and Reflective Filter Pair At
Several Cable Lengths

18

VLF-6400+ 36"

26

CONCLUSION

21

9
2900

3500

XLF-662M+ 36"

VLF-6400+ 36" @10dBm

Figure 19: Side-by-side comparisons of conversion loss ripple for a doubler-filter pair (left) and X4
multiplier chain with a filter after the first doubler (right) when conventional and reflectionless filters are used.

21

9
2900

3500

XLF-662M+ 36" @10dBm
21Frequency (MHz)Conversion Loss of Doubler and Reflective Filter Pair At
Several Power Levels

Conversion Loss (dBm)

Conversion Loss (dBm)

Conversion Loss of Doubler and Reflectionless Filter Pair At
Several Power Levels

Conversion Loss of a X4 Multiplier Chain with 2 Doublers and a
LPF either Reflective or ReflectionlessConversion Loss of A Doubler and Filter Reflective vs.
Reflectionless at 36" Cable Length

Conversion Loss (dBm)

Results of the two experiments are shown in Figures 16, 17, 18, and 19. Figure 16 shows the conversion loss of a doubler-filter
pair at various power levels with a reflectionless filter (left) and a reflective filter (right). It can be observed from the plots
in Figure 16, that even at nominal power levels for the doubler, the reflective filter’s out-of-band impedance mismatchleads to reflections of the harmonics, especially the 3rd harmonic, which leads to greater conversion loss ripple. Figure
17 exaggerates this point by replicating the experiment that produced Figure 16 with a longer transmission line length
between the filter and doubler by 6 inches, allowing for a greater development of standing waves. Figure 17 shows thatthe reflectionless filter (left) is able to effectively attenuate the out-of-band reflections with almost no additional conversion
loss ripple, while the reflective filter exhibits much worse standing wave effects the longer the transmission line used.

3000

3100

3200

3300

3400

3500

Frequency (MHz)Figure 18: Conversion loss curves for doubler-filter pairs with conventional and reflectionless filters, illustrating effects of out-of-band
impedance mismatch on standing waves in the signal chain.

Figure 18 captures the susceptibility of a doubler-filter pair to effects that can worsen the standing waves developedby filters with out-of-band impedance mismatch. Even in the worst case experimental example (a 36” transmission line
extension), the reflectionless filter (left) only develops a marginal conversion loss ripple, while the ripple of the reflective
filter (right) is over 2 dB. A side-by-side comparison of the conversion loss ripple for both the reflective and reflectionlessfilter is provided in Figure 19.

5. Morgan, M. A., & Boyd, T. A. (2015). Reflectionless Filter Structures. IEEE Transactions on Microwave Theory andTechniques, 63(4), 1263–1271. doi:10.1109/tmtt.2015.2403841
6. Morgan, M. A., Groves, W. M., & Boyd, T. A. (2018). Reflectionless Filter Topologies Supporting Arbitrary Low-Pass LadderPrototypes. IEEE Transactions on Circuits and Systems I: Regular Papers, 1–11. doi:10.1109/tcsi.2018.2872424
7. M. Morgan, “Think Outside the Band: Design and Miniaturization of Absorptive Filters,” IEEE Microwave Magazine, vol. 19, no. 7, pp. 54-62, November 2018.
8. R. Setty, B. Kaplan, M. Morgan, and T. Boyd, “Combining MMIC Reflectionless Filters to Create UWB band-pass Filters,” Microwave Journal, vol. 61, no. 3, pp. 60-72, March 2018.
9. Pairing Mixers with Reflectionless Filters to Improve System Performance, Mini-Circuits Application Note
10. Advantages of Cascading Reflectionless Filters, Mini-Circuits Application Note
11. M. Morgan, “Reflectionless Filters,” U.S. Patent No. 8,392,495, March 5, 2013. People’s Republic of China
Patent No. 201080014266.1, July 30, 2014.
12. M. Morgan, “Optimal Response Reflectionless Filters,” U.S. Patent No. 10,263,592, April 16, 2019. Taiwan Patent No. I653826, March 11, 2019.
13. M. Morgan, “Transmission Line Reflectionless Filters,” U.S. Patent No. 9,923,540, March 20, 2018, No. 10,277,189 , April 30, 2019.

28

www.minicircuits.com 29

SC SPLITTERS/COMBINERS
50Ω DC to 43500 MHz

Ultra-Wideband MMIC
Splitter/Combiners• Power handling up to 2.5W
• Low insertion loss
• Excellent isolation•N ew resistive/reactive designs extend
frequency coverage down to DC
Frequency
Range
(MHz)

Isolation(dB)

Insertion
Loss
(dB)

Phase
Unbalance(deg)

Amplitude
Unbalance(dB)

Power Input(W) asSplitter,Max.

DC Pass

EP2KA+

2

10000-43500

17

2.2

9.6

0.57

1.25

Y

EP4KA+

4

10700-31000

19.3

0.6

4.7

0.2

0.6

Y

EP2K1+

2

2000-26500

20

2.4

5.4

0.3

2.5

Y

EP2K+
40

Total Loss

2

35

5000-20000

2

EP2RKU+

EP2C+

2

15

EP2W1+
10
5

EP2RCW+0

DC-18000

0

EP2W+

1000

2000

Total Loss (dB)

1.1

20

1.8

16

2

DC-8000

23.0

4.8

12

2

3000

4000

5000

6000

700-6000

7000

FREQUENCY (MHz)

8000

19.8

9000

10000

1.3

2.5

Y

1.1

0.02

0.3

N

1.9

0.2

0.6

N

0.2

1.85

Y

2.5

Y

@+25°C

1.7

14

10

0.1

6

18

19.4

0.1
@-55°C@+105°C

0.9
0

1000

0.9

0.02

2000

3000

0.1

0.3
5000

6000

45

6.6

40

6.4

35

6.2
6.0

Total Loss S-1@-55°C5@+105°C

5.0

0

6000

8000

10000

Ultra-wideband MMIC splitter/combiners

12000

14000

16000

18000

4000

6000

8000

10000

12000

18000

8000

9000

FREQUENCY (MHz)Isolation 1-2 vs Temperature28
26

Isolation (dB)

Total Loss S-2

25

20
15

24
22
20
18

@+25°C1610@-55°C14

5@+105°C

12
0

1000

2000

3000

4000

5000

6000

7000

8000

9000

10000

10

0

1000

2000

3000

4000

5000

6000

7000

10000

FREQUENCY (MHz)FREQUENCY (MHz)

6.8

2000

EP2RCW+

Total Loss S-1

30

7.0

16000

Y

30

35

0

0

EP2RCW+

Total Loss

40

14000

2.5

25

5.2

4000

10000

20

@+25°C

2000

9000

30

10

Total Loss S-2

0

N

8000

Isolation 1-2 vs Temperature

15

5.6

7000

FREQUENCY (MHz)

50

5.8

4000

EP2RKU+

Total LossFREQUENCY (MHz)Total Loss (dB)

22

6.8

H I G H L I G H T S
▼

16

24

500-9500

5.4

▼

3.8

26

2

7.0

New resistive/reactive designs extend
frequency coverage down to DC

18.8

1800-12500

28

3.3

Isolation 1-2 vs Temperature

4.2

30

Isolation (dB)

Total Loss (dB)

20

2.1

26.1

Total Loss S-2

4

EP4RKU+
25

20

DC-18000
Total Loss
S-1

30

EP2RCW+

Isolation (dB)Model Number

Total Loss
50
45

www.minicircuits.com 31

Isolation 1-2 vs Temperature

SPLITTERS/COMBINERSSPLITTERS/COMBINERS

/
S
S
R
R
E
E
T
N
T
I
I
B
L
P
S OM
C

No. ofWays

NEW RELEASESSW SWITCHES
50Ω DC to 6000 MHz

MMIC Switches• Fast switching• Excellent isolation• With and without internal CMOS drivers• Tiny size, as small as 2x2mm
NEW RELEASESFrequency
Range
(MHz)

DriverConfiguration

Insertion
Loss
(dB)

1 dB
Compression
(dBm)

Input
IP3
(dBm)

In-OutIsolation(dB)

CSWA2-63DR+

SPDT

500-6000

CMOS

Absorb

1.1

27

45

50

VSWA2-63DR+

SPDT

500-6000

CMOS

Absorb

1.2

27

44

46

HSWA2-63DR+

SPDT

100-6000

CMOS

Absorb

1

35

65

68

HSWA4-63DR+

SP4T

30-6000

CMOS

Absorb

1.15

35

58

52

JSW2-63DR+

SPDT

5-6000

CMOS

Reflect

0.33

35

59

35

JSW2-63VHDRG+

SPDT

5-6000

CMOS

Reflect

0.36

37

75

33

JSW2-63VHDRP+

SPDT

5-6000

CMOS

Reflect

0.36

37

75

33

MSW2-50+

SPDT

DC-5000-

Reflect

0.7

23

54

53

MSWA2-50+

SPDT

DC-5000-

Absorb

0.7

24

54

53

M3SW-2-50DRA+

SPDT

DC-4500

CMOS

Reflect

0.6

25

47.3

48

M3SWA-2-50DRA+

SPDT

500-4500

CMOS

Absorb

1.2

27

46

51

M3SWA-2-50DRB+

SPDT

DC-4500

CMOS

Absorb

0.6

25.4

46.5

56

VSW2-33-10W+

SPDT

50-3000-

Reflect

0.5

40

56

26

HSWA2-30DR+

SPDT

DC-3000

CMOS

Absorb

0.9

31

55

55

HSW2-272VHDR+

SPDT

30-2700

CMOS

Reflect

0.4

45.5

85

28

JSW3-272DR+

SP3T

5-2700

CMOS

Reflect

0.6

35

59

30

JSW4-272DR+

SP4T

5-2700

CMOS

Reflect

0.6

35

59

30

JSW5-272DR+

SP5T

5-2700

CMOS

Reflect

0.6

35

59

30

JSW6-33DR+

SP6T

5-2700

CMOS

Reflect

0.6

35

59

30

MSW-2-20+

SPDT

DC-2000-

Reflect

0.5

24

-

34

MSWA-2-20+

SPDT

DC-2000-

Absorb

0.95

27

-

40

Model Number

S
E
H
C
IT
W
S

▼

H I G H L I G H T S

 ew wideband MMIC switches provideNhigh isolation and fast switching speed.

www.minicircuits.com 33

SWITCHESSPLITTERS/COMBINERS

Type

S
R
E
M
R
O
F
S
N
A
R
T

▼

LTCC Transformers▼

H I G H L I G H T SCore and Wire Transformers for CATV Systems

www.minicircuits.com 35

TS TRANSFORMERS
50Ω & 75Ω 223 to 8000 MHz

75Ω 4.5 to 3000 MHz• Tiny size, 0805
• Excellent power handling• Rugged ceramic construction• Cost effective for volume production•B roadband, supporting DOCSIS® 3.1
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Transformers for CATV

LTCC Baluns

NEW RELEASE

Single-Ended
toSingle-Ended

Single-Ended
to
Balanced

BalancedtoBalanced

CenterTap

DC
Isolation

Impedance(Ω)

Impedance
Ratio

NCS2-83+

3000-8000

N

Y

N

N

Y

50

2

NCS2-622+

5600-6200

N

Y

N

N

Y

50

2

NCS1-63+

4900-6000

N

Y

N

N

Y

50

1

NCS4-63+4500-6000

N

Y

N

N

Y

50

4

NCS2-592+

4900-5875

N

Y

N

N

Y

50

2

NCS4-442+

3300-4200

N

Y

N

N

Y

50

2

NCS1-422+

3300-4000

N

Y

N

N

Y

50

1

NCS2-392+

3000-3900

N

Y

N

N

Y

50

2

NCS1-332+

700-3300

N

Y

N

N

Y

50

1

NCS2-33+

1500-3100

N

Y

N

N

Y

50

2

NCS1-292+

1650-2850

N

Y

N

N

Y

50

1

NCS3-272+

2250-2725

N

Y

N

N

Y

50

3

Single-Ended
toSingle-Ended

Single-Ended
to
Balanced

BalancedtoBalanced

CenterTap

DC
Isolation

Impedance
Ratio

SCTX1.33-33-2W+

10-3000

N

Y

N

N

N

1.33

TC1-33-75G2+

5-3000

N

Y

Y

N

N

1

TC1-1-13M-75+

4.5-3000

N

Y

Y

N

N

1

TC1-1-13M-75X+

4.5-3000

N

Y

Y

N

N

1

TRS1-23-75+

10-2200

N

Y

Y

N

N

1

TRS1-182-75+

10-1800

N

Y

Y

N

N

1

TC1.33-182X-75+

5-1800

N

Y

N

Y

N

1.33

TCM2-142-75X+

10-1400

N

Y

Y

N

N

2

TRC1-1K122-75+

20-1250

N

Y

Y

N

N

1

TRC1-1-122-75+

5-1250

N

Y

N

N

N

1

Average Insertion Loss

3.0

NCS4-272+

2300-2700

N

Y

N

N

Y

50

4

TC1.33-182X-75+
35

Y

N

N

Y

50

4

NCS2-232+900-2300

N

Y

N

N

Y

50

2

2.4
2.2
2.0
1.8

NCS1.5-232+

400-2300

N

Y

N

N

Y

50

1.5

NCS2-222+

1275-2200

N

Y

N

N

Y

50

2

1.4

2

1.0

NCS2-222-2+

1275-2200

N

Y

N

N

Y

50

1.6

N

Y

N

N

Y

75

1

NCS1-23+

1300-2000

N

Y

N

N

Y

50

1

NCS1-112+

700-1100

N

Y

N

N

Y

50

1

2.0

NCS4-102+

700-1000

N

Y

N

N

Y

50

4

1.6

N

Y

N

N

Y

50

2

NCS2-771-75+

240-770

N

Y

N

N

Y

75

2

NCS3-72+

250-760

N

Y

N

N

Y

50

3

NCS2-62+390-590

N

Y

N

N

Y

50

2

NCS1-521+

223-520

N

Y

N

N

N

50

1

NCS4-521+

223-520

N

Y

N

N

Y

50

4

25
20
15
10

0

200

400

600

800

1000

1200

1400

1600

1800

0

2000

0

200

400

600

Average Insertion Loss

TRC1-1K122-75+

1000

1200

1400

1600

1800

2000

1050

1200

1350

1500

Input Return Loss

40

1.8

35

1.4
1.2
1.0
0.8
0.6
0.4

30
25
20
15
10
5

0.2
0.0

800

Frequency (MHz)Frequency (MHz)950-2200

240-770

30

5

1.2

NCS1-222-75+

NCS2-771+

Input Return Loss (dB)N

Input Return Loss (dB)

1600-2300

Insertion Loss (dB)

2.6

NCS4-232+

Input Return Loss

40

2.8

Insertion Loss (dB)

TRANSFORMERSFrequency
Range
(MHz)

Model
Number

0

150

300

450

600

750

900

Frequency (MHz)

1050

1200

1350

1500

0

0

150

300

450

600

750

900

Frequency (MHz)www.minicircuits.com 37

TRANSFORMERSSPLITTERS/COMBINERS

Frequency
Range
(MHz)

Model
Number36

NEW RELEASESINTRODUCTION

DEMYSTIFYING RF
TRANSFORMERSPart 1: A Primer on Transformer Theory,
Technologies, and Applications

William Yu and Urvashi Sengal, Mini-Circuits

In essence, a transformer is merely two or more conductive paths linked by a mutual magnetic field. When a varying
magnetic flux is developed within the core by alternating current passing through one conductive path, a current isthen induced in the other conductive paths. This induced current is proportional to the ratio of magnetic coupling
between the two conductive paths. The ratio of the magnetic coupling of the conductive paths with the core
determines the induced voltage in the additional conductive paths, providing both an impedance transformation
and a voltage step-up or step-down. Additional conductive paths, potentially all with different coupling ratios, may
be added to realize various functions, which is why RF transformers are such varied and versatile devices used widely
throughout the RF/microwave industry.
A common implementation of an RF transformer consists of two or more distinct wires wrapped around a magnetic core (or air
core for higher frequencies), which is why RF transformers are often described as a ratio of the number of windings, or turns.
RF transformers are used for a wide variety of applications, as the nature of the device allows for various
configurations that can serve a variety of different functions, including (but not limited to):• Injecting DC current• Providing DC isolation between circuits• Enhancing common mode rejection• Providing an impedance transformation for impedance matching• Efficiently coupling between balanced and unbalanced circuits• Inducing a voltage/current step-up or step-down
There are several common technologies used to build transformers, including core-and-wire, transmission line, lowtemperature co-fired ceramic (LTCC), and monolithic microwave integrated circuit (MMIC). Each comes in a variety of
packages and with a wide range of performance characteristics.

TRANSFORMER THEORY AND PRACTICAL INSIGHTS

Though not viable for practical applications, the basic Ideal Transformer Model is helpful in illustrating thefundamental behavior of transformers. Consider the diagram, Figure 1. The current through the primary winding
creates a magnetic flux (Faraday’s Law) through the mutual magnetic field of the core, inducing a proportional current
and voltage in the secondary winding. Both the current and voltage developed are proportional to the windings ratio,or magnetic coupling between the windings and the core.

(Fig 1)Hence, the secondary impedance is a function of the square of the windings ratio times the impedance of the primary(Equation 1). Where I1, V1, and Z1 are the current, voltage, and impedance through the primary winding; I2, V2, andZ2 are the current, voltage, and impedance through the secondary winding, N1 is the number of turns in the primarywinding; N2 is the number of turns in the secondary winding; 1 and 2 are the input ports of the primary winding; andlastly, 3 and 4 are the output ports of the secondary winding.

However, a real transformer intrinsically includes several parasitic resistances, inductances, and capacitances, of which
there are both mutual and self-parasitic capacitances. Illustrated in Figure 2 is a lumped-element model of a non-idealtransformer, which depicts parasitic resistances and inductances of the two windings, as well as core resistive losses
and the windings’ active inductance. It is clear from the model for a non-ideal transformer that these devices operatewith a limited bandwidth, have insertion loss, a maximum power rating, and exhibit other frequency-, temperatureand power-dependent performance.

38

www.minicircuits.com 39

TRANSFORMERSThis article is the first part in our Demystifying RF Transformers series, and focuses on introducing RF transformer
theory with discussion of common RF transformer technologies and applications.

DEMYSTIFYING RF TRANSFORMERS: P1

CORE-AND-WIRE RF TRANSFORMERS(Fig 2)

Typically, a real RF transformer’s lower cutoff frequency is dictated by the winding’s active inductance, and the highfrequency cutoff is dominated by the inter-winding and intra-winding capacitance. Insertion loss present in the operating
bandwidth is a product of the primary and secondary winding ohmic losses, as well as the dissipation within the core. As
ohmic losses tend to be a function of frequency and temperature, the effective operating bandwidth of a transformer islimited by these factors. Several RF transformer types also introduce leakage inductances due to incomplete magnetic
coupling between the windings. As the leakage inductance’s reactance is proportional to frequency, these parasitics
reduce the return loss at high frequencies but increase the insertion loss at lower frequencies.

TRANSFORMERS

A common Balun fashioned from a transformer is a flux coupled Balun Transformer, which is constructed by winding
separate wires around a magnetic core and grounding one side of the primary winding. Hence, the single-ended
RF signals entering the primary unbalanced winding side undergo an impedance transformation to a differential(balanced) output through the secondary winding. A deeper examination of Balun and Unun theory, functions, andperformance will be presented in Part 2 of the Demystifying RF Transformers series.

TRANSMISSION LINE RF TRANSFORMERSThere are a variety of transmission line transformer topologies, which may include precisely designed transmission
lines placed between two mismatched loads, or a complex arrangement of several transmission lines. For instance, alength of transmission line can be used to implement an impedance transformation between two mismatched loads.
Some transmission line transformers use insulated wires wrapped around ferrite cores, which closely resemble typicalcore-and-wire transformers, and are often considered core-and-wire transformers. The following explanation is less for
categorization purposes, but meant to describe aspects of transformer behavior and enhance understanding.

(Fig 5) Ideal transmission line transformer

A basic transmission line transformer consists of a two-conductor transmission line. The first conductor is connectedfrom the generator to the load, and the other conductor is connected at the output of the first transmission line andthe load to ground, as seen in Figure 6. With this configuration, the current flowing through the load is twice thecurrent flowing through the generator, and V0 is one half the voltage V1. Hence, the load resistance is only a quarterof the resistance seen at the generator side, yielding a 1:4 transformer (Equation 2).
A common transmission line transformer variation is the quarter-wave transmission line transformer. This topology
uses a transmission line with a characteristic impedance that enables impedance matching between the input
impedance and the load. The length of a quarter-wave transformer is dictated by the desired frequency of operation,and is limited to one bandwidth octave around the center frequency. Consider a lossless transmission line with Z0
characteristic impedance and length L, connected between Zin input impedance and ZL load impedance as seenin Figure 7. In order to match Zin with ZL, the characteristic impedance of the quarter-wave transmission line would
need to be the square root of Zin multiplied by ZL (Equation 3).

(Fig 3)(Fig 4)NOTES ON TRANSFORMER MAGNETIC CORES

RF transformers that include a magnetic core (typically ferromagnets) are subject to several undesirable
performance degrading factors. For instance, the magnetizing inductance of the core limits the low frequency
performance of an RF transformer. The magnetizing inductance is a function of the core permeability, crosssectional area of the magnetic core, and the number of windings around the core. The magnetizing inductanceincreases the insertion loss at low frequencies and also decreases the return loss. Furthermore, the permeabilityof the core is a function of temperature, for which increases in permeability subsequently increase the low
frequency insertion loss.

RF TRANSFORMERS TECHNOLOGIESThe two main types of discrete RF transformers are core-and-wire and transmission line transformers. Additionally,
there are two common types of low-profile and compact RF transformer designs: LTCC and MMIC transformers.

40(Fig 6)(Eq 2)

An advantage of a transmission line transformer is that a significant portion of the interwinding capacitance isassumed by the transmission line parameters, along with the leakage inductance, which results in a comparatively
wider operating bandwidth compared to core-and-wire transformers.

www.minicircuits.com 41

TRANSFORMERS

More complex RF transformer topologies, i.e. transformers with several windings, taps, and additional elements
present varying performance dynamics based on the topology and transformer construction. For example, an RF
device, known as a Balun, which is used to efficiently interconnect balanced (differential signal path) RF circuits
to unbalanced (single-ended signal path) RF circuits using impedance transformation, can be realized with an
RF transformer. Moreover, another device similar to a Balun, known as an Unun, which is used to interconnect
unbalanced to unbalanced RF circuits, can also be realized with an RF transformer.

Core-and-wire RF transformers are fabricated by wrapping conductive wires, typically insulated copper wires, arounda magnetic core, such as a toroid. There may be one or more secondary windings, which may also be center tapped
to enable additional functions. Figure 5 shows an RF transformer made from a toroidal magnetic core and insulated
copper windings. Due to the nature of the inductive coupling between the wires and the core, smaller core and wiredimensions tend to yield core-and-wire transformers that can operate at much higher frequencies than larger
core-and-wire transformers. However, the smaller dimensions of compact core-and-wire transformers also increase
the resistive losses of the windings and the core, which exhibits greater insertion loss at lower frequencies.

DEMYSTIFYING RF TRANSFORMERS: P1(Fig 7)(Fig10)(Eq 3)

Low Temperature Co-fired Ceramic (LTCC) components are multilayer components that are
fabricated using a ceramic-based substrate. LTCC transformers use coupled lines that act
as transmission lines in order to achieve impedance transformation and signal conversion
from single-ended to balanced. LTCC transformers rely on capacitive coupling to operate.
This allows LTCC transformers to operate at higher frequencies compared to ferromagnetictransformers, but may lead to performance degradation at low frequencies. The benefit of
LTCC technology is the ability to fabricate small and rugged transformers ideal for hi-rel
applications, as seen in Figure 8.

(Fig8)MMIC TRANSFORMERS

Like LTCC technologies, MMIC transformers are made using high-precision
layered 2D substrates with planar metallization. Typically, MMIC transformers arefabricated using spiral inductors printed on a substrate in a two transmission line
configuration parallel to each other. The MMIC process is helpful for producing
high frequency transformers, and exhibits outstanding repeatability and excellent
thermal efficiency. A MMIC transformer can be fabricated using gallium arsenide(GaAs) integrated passive device (IPD) process, as seen in Figure 9.

(Fig11)(Fig9)INTRODUCTION TO RF TRANSFORMER FUNCTIONS & APPLICATIONS

As indicated in previous sections of this article, there are a variety of functions that RF transformers can performdepending upon their typology.

Transformation Functions

For instance, an impedance transformation function can be used to match two circuits with disparate 		
impedance, or also provide a voltage step-up or step-down respective to abalanced (differential) circuits.

RF Choke

Other topologies can be used to filter high frequency RF components from a signal line, functioning as a choke.

CONCLUSION

Impedance Matching Transformers

RF transformers can be fabricated with a wide variety of manufacturing methods, with a diverse range of materials,
and can be configured into a myriad of topologies in order to perform useful functions for RF circuits. Dependingon the materials, construction, and design, RF transformers can be narrow- or wide-band, and operate at low RF
frequencies or high. Understanding the basics, as well as more detailed nuances, of RF transformers can help adesigner optimize a design with the right transformer selection.

Ununs and Baluns

This concludes Part 1 of the Demystifying RF Transformers Series. Stay tuned for the following articles:
Demystifing RF Transformers Part 2: Baluns & Ununs
Demystifing RF Transformers Part 3: Understanding RF Transformer Performance Parameters
Demystifing RF Transformers Part 4: How to Select the Ideal Transformer

There are many cases in RF circuits where a mismatch in impedances between two nodes can cause reduced
power transfer efficiency and potentially troublesome reflections. Hence, an impedance matching 		transformer can be used to effectively eliminate the reflections and provide maximum power transfer 		between the two circuit nodes (see Figure 10).
Moreover, for unbalanced lines, an autotransformer configuration can be used for impedance matching, 		
i.e. an Unun. As mentioned earlier, Balun transformers can also be used to interconnect balanced 		and unbalanced circuit sections.

DC Isolation

An RF transformer can also be designed to provide DC isolation between the primary and 			secondary windings, which can be useful for separating RF circuits that require a DC bias on the RF 		transmission line from RF circuits that would be negatively impacted by a DC voltage on the signal line.

42

Enhancing Common Mode Rejection
Certain RF transformer designs can be used to provide enhanced common mode rejection (CMR) for	

RESOURCES

1. https://www.minicircuits.com/app/AN20-001.pdf
2. https://www.minicircuits.com/app/AN20-002.pdf
3. https://www.minicircuits.com/appdoc/TRAN14-2.html
4. RF and Microwave Transformer Fundamentals featured in Microwave Products Digest 10-2009
5. https://www.minicircuits.com/WebStore/Transformers.html

www.minicircuits.com 43

TRANSFORMERSTRANSFORMERSLTCC TRANSFORMERSInjecting DC Current

Moreover, if a DC current is required for a subsection of a circuit, a specialized RF transformer can be used
to inject DC current into the signal path. For example, two center tapped transformers can be used to injecta DC bias, replacing the need for two Bias Ts (see Figure 11).

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H I G H L I G H T S

LTCC Designer’s Kits for Wi-Fi,
Bluetooth and Zigbee
To support customers working in Wi-Fi,
Bluetooth and Zigbee applications,Mini-Circuits has developed a series of ourpopular LTCC filters and baluns optimized forthe 2400 to 2500 MHz and 4900 to 5900 MHz
bands. Two new designer’s kits provide aselection of filters, diplexers, couplers andbaluns for evaluation and prototyping.

www.minicircuits.com 45

DK DESIGNER'S KITS
2.4 to 2.5 GHz / 4.9 to 5.9 GHz

2.4 to 2.5 GHz / 4.9 to 5.9 GHz

FILTERS / DIPLEXERS / BALUNSCOUPLERS / BALUNS

K1-LTCC-WBZ+

K2-LTCC-WBZ+• Perfectly suited for WiFi, Bluetooth & Zigbee applications• Incredibly small sizes down to 0202
• LTCC robust construction• Excellent Power Handling, Up to 2 Watts• Low cost

• Perfectly suited for WiFi, Bluetooth & Zigbee applications• Incredibly small sizes down to 0402
• Rugged LTCC construction• Excellent Power Handling• Low cost

K2-LTCC-WBZ+ Electrical Specifications (21 models, 5 of each, 105 Total)

K1-LTCC-WBZ+ Electrical Specifications (20 models, 5 of each, 100 Total)

4900-5900

LPJC-592R+

DIPLEXERS

Model
NumberFrequency
Range
(MHz)

Impedance
Low(Ω)

Impedance
Ratio

BLNK1-542R+

4900-5950

501

BLJC1-542R+

4900-5950

50

1

40

BLNK2-542R+

4900-5950

5027200-7500

34

BLGE1-542R+

4900-5875

50

1

40
37

BLJC2-542R+

4900-5875

502

BLGE2-542R+

4900-5875

50

2

(MHz)

Rejection
Typ. (dB)(MHz)

Rejection
Typ. (dB)

0.7

9800-11800

4214700-17700

54

4900-5950

0.4

8800-12600

49

LPNK-252R+

2400-2500

0.5

4800-500042

LPJC-252R+

2400-2500

0.5

4800-5000

52

7200-7500

LPGE-252R+

2400-2500

0.3

4800-5000

44

7200-7500
9600-10000

HPSC-492R+

4900-5900

0.4

500-2400

25

2400-2500

34

HPJC-492R+

4900-5850

0.5500-2400

25

2400-2500

36

HPJC-252R+

2400-2500

1.0

500-1917

35

1917

25
32
38

BPNK-542R+

4900-5950

1.3

2400-2500

23

9800-11900
14700-17850

BPJC-542R+

4900-5900

1.0

DC-270040

9800-12000

34

BPGE-542R+

4900-5920

0.9

3500

49

9800-11840
14700-17760

32
30

3.3

695-800
1910

40
25

3200
4800-5000
7200-7500

36
23
22

46
26

3200
4800-5000
7200-7500

38
29
43

42
14

3000
3600-3800
4800-5000

18
43
37

BPNK-252R+

2400-2500

BPJC-252R+

2400-2500

2.3

695-800
1910

BPGE-252R+

2400-2500

1.3

1200-1300
2000

BFGE2-552R+
BFGE1-252R+

4800-5875
2400-2500
2400-2500

1.3

3500

49

1.9

DC-1000
1000-2000

44
39

4800-5000
7200-7500

49
34

1.7

880-960
1710-1910

59
38

4800-5000
7200-7500

42
32

30
40
15
24
30

4800-6000
7200-7500

2622

DPNK-252-492R+

2400-2500
5150-5850

0.4
1.2

700-2025
2400-2690
3500-3700
7250-7800
10300-11700

DPJC-252-492R+

2400-2500
4900-5950

0.7
0.7

800-2500
9800-11900

34
37

4800-6000
7200-7500

28
28DPGE-252-492R+

2400-2500
4900-5950

0.4
0.5

800-2500
9800-11900

25
194800-6000
7200-7500

36
33

BALUNSLOW PASSFILTERSHIGH PASSFILTERS
BAND PASSFILTERSBALANCEDFILTERS

LPGE-592R+

Additional Stopbands

Case Styles: AT224-1, AT1521,
BLJC4-542R+
4900-5875
50
DB1627, GU2644

DESIGNER'S KITS
SPLITTERS/COMBINERS

DESIGNER'S KITS

(MHz)

Ins. Loss
Typ. (dB)

BFNL2-252R+

46

Stopband 1

COUPLERS

Passband
Model
Number4

BLGE4-542R+

4900-5875

50

4

BLNK1-252R

2400-2500

501

BLJC1-252R

2400-2500

50

1

BLGE1-252R+

2400-2500

501

BLNK2-252R+

2400-2500

502

BLJC2-252R+

2400-2500

502

BLGE2-252R+

2400-2500

502

BLJC4-252R+

2400-2500

50

4

BLGE4-252R+

2400-2500

50

4

Model
NumberFrequency
Range
(MHz)

Coupling
Nom.
(dB)

Mainline
Loss (dB)Typ.

Directivity
(dB)
Typ.

VSWR
(:1)
Typ.

Power
Input
Max. (W)

CPJC-6-252R+

2400-2500

6.5

1.27

18

1.2

2

CPJC-10-252R+

2400-2500

10

0.65

19

1.33

2

CPJC-17-252R+

2400-2500

17.65

0.14

12

1.06

2

CPJC-21-252R+

2400-2500

21

0.3

19

1.1

2

CPJC-28-252R+

2400-2500

28

0.3

10

1.05

2

www.minicircuits.com 47

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