Beyond 5G White Paper
Supplementary Volume“Radio Technologies for higher frequency”

Version 1.0
March 7, 2024
Beyond 5G Promotion Consortium
White Paper Subcommittee

Preface .............................................................................................................................. 6
I. Spatial Characteristics of Sub-THz Band Radio Propagation in Indoor Environments
for Beyond 5G........................................................................................................... 9
I-1. Introduction .............................................................................................................. 9I-2. Measurement Environment and Measurement Method ..................................... 10
I-3. Measurement Results ............................................................................................ 11
I-4. Conclusion .............................................................................................................. 13
II. Path Loss Characteristics from Microwave to Sub-THz Bands in Urban Environment
for Beyond 5G......................................................................................................... 14
II-1. Introduction .......................................................................................................... 14
II-2. Key Propagation Phenomena Affecting Radio Channel Characteristics in bands
above 100 GHz...................................................................................................... 14
II-3. Measurement System Setup ................................................................................ 15
II-4. Path Loss Characteristics .................................................................................... 15
II-5. Building Penetration Loss ................................................................................... 17
II-6. Conclusion ............................................................................................................. 18
III. Evaluation Technique for Propagation Characteristics of 300-GHz-band Radio wave
near the Human Body ............................................................................................ 20
III-1. Introduction ......................................................................................................... 20
III-2. Shadowing loss due to human body ................................................................... 20
III-3. 300 GHz band human body phantom ................................................................ 21
III-4. DOA measurement result ................................................................................... 22
III-5. Conclusion............................................................................................................ 24
III-6. Acknowledgements.............................................................................................. 24
IV. AI Calibration Technique for RF Impairments with Sub-THz transmission ......... 25
IV-1. Introduction ......................................................................................................... 25
IV-2. Concept of AI calibration .................................................................................... 26
IV-3. Compensation technique based on AI ................................................................ 28
IV-4. Conclusion ............................................................................................................ 29
V. Sub-THz-Band Massive MIMO Technology for Beyond 5G/6G ................................ 31
V-1. Introduction ........................................................................................................... 31

1

V-2. M-MIMO Technology Based on Hybrid BF ......................................................... 31
V-3. Performance Evaluation of 100 GHz-Band M-MIMO by Link-Level Simulations............................................................................................................................... 32
V-4. Simulation Results of 100 GHz-Band M-MIMO Technology............................. 33
V-5. Conclusion ............................................................................................................. 35
VI. High-Capacity Wireless Transmission Technology using Orbital AngularMomentum Multiplexing in the Sub-THz Frequency Bands ................................ 37
VI-1. Introduction ......................................................................................................... 37
VI-2. OAM Multiplexing Transmission and Use Cases ............................................. 37
VI-3. Enabler of OAM Multiplexing in Sub-THz Frequency Bands ......................... 38
VI-4. Experimental Evaluations .................................................................................. 40
VI-5. Conclusion ............................................................................................................ 41
VII. Performance Analysis of OAM Multiplexing Using Parabolic Antenna ................ 43
VII-1. Introduction ........................................................................................................ 43
VII-2. Performance Analysis of OAM Multiplexing Using Parabolic Antenna ........ 44VII-3. Numerical Results ............................................................................................. 45
VII-4. Conclusion .......................................................................................................... 47
VIII. Antenna Array Module for Sub-THz Communications ........................................ 48
VIII-1. Introduction ...................................................................................................... 48
VIII-2. Demonstration of 1 × 4 Antenna Array .......................................................... 49
VIII-3. Conclusion ......................................................................................................... 52
VIII-4. Acknowledgements ........................................................................................... 52
IX. Virtualized terminal technologies for high-capacity communications in Beyond 5G
era .......................................................................................................................... 53
IX-1. Introduction ......................................................................................................... 53
IX-2. Virtualized terminal technologies ...................................................................... 53
IX-3. Demonstration experiment ................................................................................. 55
IX-4. Conclusion ............................................................................................................ 57
IX-5. Acknowledgements .............................................................................................. 57
X. Sub-THz multi-beam antennas for virtualized terminal technologies ..................... 58
X-1. Introduction ........................................................................................................... 582

X-2. Required function of Sub-THz antennas for virtualized terminal technologies............................................................................................................................... 58
X-3. Plano-convex dielectric lens multi-beam antennas ............................................ 59
X-4. Spherical dielectric lens multi-beam antennas .................................................. 60
X-5. Rotman lens multi-beam antennas ..................................................................... 61
X-6. Conclusion ............................................................................................................. 61
X-7. Acknowledgements ............................................................................................... 62
XI. Sub-THz band device technologies ........................................................................... 63
XI-1. Introduction ......................................................................................................... 63
XI-2. Frequency Domain Equalization Technology .................................................... 64
XI-3. Non-Linear Components Compensation Technology ........................................ 65
XI-4. Device technology for 100 GHz band power amplifier ...................................... 65
XI-5. Device technology for 300 GHz band power amplifier ...................................... 66
XI-6. Challenge of utilizing over 100 Gbit/s per channel ........................................... 68
XI-7. Beamforming in 300-GHz Band ......................................................................... 69
XI-8. Conclusion ............................................................................................................ 70
XI-9. Acknowledgements .............................................................................................. 70
XII. 6G Simulator using Real Environment Model based on Point Cloud Data ........... 72
XII-1. Introduction ........................................................................................................ 72
XII-2. Overview of 6G Simulator Based on a Real Environment Model Using Point
Cloud ..................................................................................................................... 73
XII-3. Simulation Results ............................................................................................. 75
XII-4. Conclusion .......................................................................................................... 76
XIII. THz Sensing and Communication Towards Future Intelligence Connected
Networks ................................................................................................................ 78
XIII-1. Introduction ...................................................................................................... 78
XIII-2. THz Spectrum and Application Scenarios ...................................................... 78
XIII-3. THz Channel Propagation and Modeling ....................................................... 78
XIII-4. THz Hardware and Components ..................................................................... 79
XIII-5. THz System and Testing .................................................................................. 81
XIII-6. Conclusion ......................................................................................................... 843

XIV. 300GHz point-to-point fixed wireless transmission systems ................................ 86
XIV-1. Introduction ....................................................................................................... 86
XIV-2. 300GHz Fixed wireless transmission using V-band modems........................ 87
XIV-3. THz-wave scattering ......................................................................................... 90
XIV-4. Conclusion ......................................................................................................... 91
XIV-5. Acknowledgements ........................................................................................... 91
Abbreviation List ........................................................................................................... 944

【Revision History】
Ver.
Date
1.0 2024.3.7
Initial version

Contents

5NotePreface
In our published white paper “Beyond 5G White Paper ~Message to the 2030s~”, weresearched what are required in various industries in the Beyond 5G era, then we
proposed “Further enhancement of specific 5G features” such as ultra-high speed
wireless access at 10 times faster than 5G’s, 1/10th of 5G’s latency. To address thesetarget, we need possibility to use a drastically wider bandwidth compared with 5G. The
higher frequency spectrum, spanning from millimeter waves to THz waves, unlocks thepotential for significantly wider bandwidth. This leap enables data transfer rates
exceeding 100 Gbps and process massive amounts of data from any location instantly
and accurately. Moreover, the short wavelength of the THz spectrum allows for theminiaturization of antenna elements, leading to the development of ultra-massive MIMO
systems. These systems promise enhanced coverage through pencil-beamforming andheightened spectrum efficiency via superior spatial resolution and frequency reuse.
However, the path to integrating THz technology is not without its challenges.
Terahertz waves tend to travel through a straight path than millimeter waves andcannot propagate over a long distance. In order to address this problem, it is necessary
to carry out technical examination on terahertz waves to clarify their radio propagation
characteristics, establish their propagation model and high-precision propagation
simulation technique, and advance device technology. A wide range of frequency
utilization technologies, such as spectrum sharing, and a review of spectrum utilization
methods in low frequency bands, will also be important.
In addition to the extremely high data rates, high-resolution sensing and imaging, andhigh-precision positioning applications will be implemented into THz communications,whose THz spectrum has features such as wider bandwidth operation, higher gain
antenna performance, and frequency-selective resonant absorption characteristics.
Moreover, as a further coverage extension technique, RIS reflectors can be applied on
the wall of the room or the building to deliver THz waves to the non-covered areas. These
technologies, ultra-massive MIMO, RIS and Sensing technologies, are discussed in detailin our other white papers.
To tackle these challenges and to realize new use cases, a lot of research anddevelopment activities in Japan. In this white paper, we show these research anddevelopment activities and its results with a lot of figures as follows.
-“Spatial Characteristics of Sub-THz Band Radio Propagation in Indoor
Environments for Beyond 5G” describes they measured and analyzed path loss
and direction of arrival of multipath in sub-terahertz wave bands of 160 GHz and
300 GHz in an indoor conference room.

6-"Path Loss Characteristics from Microwave to Sub-THz Bands in Urban
Environment for Beyond 5G” reports the path loss, namely basic transmission loss,
and building penetration loss from microwave to sub-THz bands in urban
environment to exploit the new spectrums for beyond 5G.

-“Evaluation Technique for Propagation Characteristics of 300-GHz-band Radio
wave near the Human Body” describes the evaluation techniques for 300-GHz
radio propagation near the human body, including diffraction and reflection
characteristics near the human body.

-“AI Calibration Technique for RF Impairments with Sub-THz transmission”
presents the conceptual overview of artificial intelligence (AI) device calibration
and AI calibration network to optimize communication quality and calibration
cost based on AI. Furthermore, as a basic examination of AI device calibration,
they demonstrate a compensation technology of RF impairments based on deepneural network (DNN).

-“Sub-THz-Band Massive MIMO Technology for Beyond 5G/6G” introduces linklevel simulations of a 100 GHz-band radio access technology exploiting Massive
MIMO, over-100 Gbps downlink throughput and coverage performances areclarified in the sub-THz band.

-“High-Capacity Wireless Transmission Technology using Orbital AngularMomentum Multiplexing in the Sub-THz Frequency Bands” describes they
designed and implemented an 8×8 Butler matrix that performs discrete Fouriertransform (DFT) processing in an analog way to generate and separate OAM
waves with high precision over 32 GHz bandwidth and a parabolic reflector
antenna to extend the transmission distance.

-"Performance Analysis of OAM Multiplexing Using Parabolic Antenna” describesthat they investigate OAM multiplexing transmission with a parabolic antenna
for a link distance and analyze the received power concerning link distance,
carrier frequency, and receiving antenna radius, as well as highlight theadvantages of planar reception achieved by the parabolic antenna. Additionally,
they also demonstrate how OAM modes affect propagation attenuation at
different link distances.

-“Antenna

Array

Modulefor

Sub-THz

Communications”

reports theirdevelopments and achievements of an antenna array module with element pitch
less than 1 wavelength (1λ) at 300 GHz.

-

“Virtualized terminal technologies for high-capacity communications in Beyond
5G era” presents a "virtualized terminal" concept for enhanced uplink capacity inbeyond 5G networks to overcome the space and power constraints of conventional
MIMO systems, and also reports their experimental results.

7-"Sub-THz multi-beam antennas for virtualized terminal technologies” describes
their development of multi-beam antennas using Plano-convex lenses, Sphericallenses and Rotman lens arrays at Sub-THz band to apply to the virtualized
terminal technologies of user-centric architecture for future beyond 5G/6G
systems.

-“Sub-THz band device technologies” presents sub-THz band device technologies
from high power transistors to RF components with the experiments andsimulation results.

-“6G Simulator using Real Environment Model based on Point Cloud Data”
describe an overview of the functions of the 6G system level simulator they havedeveloped and an example of its performance evaluation.

-“THz Sensing and Communication Towards Future Intelligence Connected
Networks” describes their research and prototype verification of ISAC feature on
THx spectrum.

-“300GHz point-to-point fixed wireless transmission systems” describes a 300 GHz
fixed radio transmission system based on V-band IF signal processing and also
recent research results on reflection loss and scattering due to roughness.

In conclusion, as we embark on the journey towards Beyond 5G technologies, the THz
spectrum emerges as a pivotal element in this technological evolution. Japan's endeavor
to overcome the challenges posed by THz wave usage, coupled with its commitment toresearch and development in this domain, positions it at the forefront of this next
generation communication revolution. This white paper aims to provide a comprehensive
overview of the potential, challenges, and future directions of THz technology in the
Beyond 5G landscape, with a particular emphasis on its implications and advancements
within Japan.
This White Paper was prepared with the generous support of many people who
participated in the White Paper Subcommittee. The cooperation of telecommunicationsindustry players and academia experts, as well as representatives of various industries
other than the communications industry has also been substantial. Thanks to everyone’s
participation and support, this White Paper was able to cover a lot of useful information
for future business creation discussions between the industry, academia, andgovernment, and for investigating solutions to social issues, not only in thetelecommunications industry, but also across all industries. We hope that this White
Paper will help Japan create a better future for society and promote significant global
activities.
Kazunori Sakumoto, Fujitsu

8

I. Spatial Characteristics of Sub-THz Band Radio Propagation in Indoor
Environments for Beyond 5G
Nobuaki Kuno, Koshiro Kitao, Takahiro Tomie, Satoshi Suyama
NTT DOCOMO, INC.
Mitsuki Nakamura, Minoru Inomata, Wataru Yamada
NTT Access Network Service Systems Laboratories, NTT Corporation.

Abstract— For the 6th generation mobile communication system (6G), realization ofultra-high speed and large capacity communication over 100 Gbps is examined. To
achieve communications over 100 Gbps, one method is using sub-terahertz bands from
100 GHz to 300 GHz which provide ultra-wide bandwidth. However, because of large
propagation loss, these band are suitable for short communication range, and indoor
environment will be an important scenario. In indoor environment due to large blocking
loss of obstacle, it is necessary to investigate methods to reduce the effects of blocking,such as utilizing reflected waves from walls and ceilings. Therefore, it is necessary to
clarify the path loss of direct and reflected waves and the direction of arrival waves. In
this paper, we measured and analyzed path loss and direction of arrival of multipath in
sub-terahertz wave bands of 160 GHz and 300 GHz in an indoor conference room.
I-1. Introduction
The actualization of ultra-high-speed and high-capacity communications exceeding
100 Gbps is under investigation for the 6th generation mobile communication system(6G) [1]. In order to actualize a communications speed of greater than 100 Gbps, the useof the sub-terahertz wave band from 100 GHz to 300 GHz, which can utilize a wide
frequency bandwidth, is attracting attention. Various studies on sub-terahertz charged
wave propagation have been reported [2] - [4].
Various organizations are investigating sub-terahertz bands for 6G. NTT DOCOMO
has published a 6G white paper, which indicates a frequency band of up to 300 GHz as
the scope of a 6G study [1]. This white paper shows that the communications in the 6G
era can be carried out in various environments at ultra-high speeds and with highcapacity. In order to achieve this, it is necessary to investigate the propagation
characteristics of sub-terahertz bands in various environments.
The path loss of sub-terahertz wave band is larger than that of millimeter wave.
Therefore, the indoor line-of-sight environment, where path loss is relatively smaller
than the outdoor or the non-line-of-sight environment, is considered to be a leading
environment where sub-terahertz waves are used. For system evaluation, a channel
model

for

standardization

such

as

International

9

Telecommunication

Union

Radiocommunication Sector (ITU-R) M. 2412 is necessary, and for model construction,clarification of path loss, delay and angular characteristics is necessary [5].
This paper reports the results of measurements of arrival waves from walls and theceiling of indoor conference rooms using two frequencies, 160 GHz and 300 GHz, inindoor environments. This paper describes the measurement environment andmeasurement method using 160 GHz and 300 GHz indoors. Next, this paper presents
the measurement results of the path loss of the arrival wave from a direction other than
the Tx antenna direction, which is achieved when changing the direction of a highdirectivity Tx antenna and examining the strength and direction of the arrival wave.
I-2. Measurement Environment and Measurement Method
Measurements were taken in a conference room in the NTT DOCOMO R&D Center.
Fig. I-1 shows the measurement environment and Table I-1 gives the measurement
parameters. As shown in Fig. I-1(a) and Fig. I-1(b), the measurement environment is aspace surrounded by walls and glass windows on all sides. In the diagram in Fig. I-1(a),the right side is a wall in which multiple partitions are arranged in parallel. At thebottom of the figure there is a wall and an entrance door. The left side is a wall, and apart of the wall near the top of the figure contains a glass window. The wall at the top of
the figure has a single glass window. The size of the room is approximately 12 m square,the height of the ceiling is approximately 3.6 m, and desks and chairs are arranged suchthat they surround the center of the room. The height of the desks is approximately 70
cm, and the height of the chairs is approximately 83 cm. The desks have metal legs. The
chairs have metal legs and a plastic back.
For the measurement, two frequencies with the center frequency of 160 GHz and 300
GHz are used. The Tx antenna is placed in the corner of the room, and the Rx antennais placed at measurement point A, which is at the center of the room, or measurement
point B, which is set at a point in the room diagonal to the Tx antenna. To acquire thewaves reflected from the wall surface, the horizontal direction of the Tx antenna ischanged every 10 degrees from the 0-degree direction to the 90-degree direction in Fig.
I-1(a). The ceiling direction is set to + for the angle of elevation and the ground directionto – for the angle of elevation, and the angle of elevation direction is changed every 10
degrees from -30 degrees to 30 degrees. The Rx antenna is rotated horizontally 360
degrees in the direction of each Tx antenna, and the angle of elevation direction ischanged every 3 degrees from -12 degrees to 30 degrees. From the received level obtainedby measurement, the path loss is calculated based on the transmitter used for themeasurement and the Tx and Rx antenna gain.

10

Glass window

1m

7.48 mA

Wall

14.95 mWall

11.8 m11.34 mGlass window

Tx
1m

6.4m

6.17 m
B

270°
180°

1m 1m

0°

90°
Entrance

Wall
Entrance(a) Diagram of measurement environment(b) Photo of measurement environment

Fig. I-1 Measurement environment.
Table I-1. Measurement parameters
Parameter
Value
Center frequency
160 GHz
300 GHz
Transmission signal
CW
Tx antenna
Horn antenna
Rx antenna
Cassegrain antennaTx azimuth HPBW
18 deg.
10 deg.
Tx elevation HPBW
17 deg.
9 deg.
Rx azimuth HPBW
0.9 deg.
0.9 deg.
Tx elevation HPBW
0.9 deg.
0.9 deg.
Tx antenna height3m
Rx antenna height1m
I-3. Measurement Results
Fig. I-2 (a) shows the minimum value of the path loss in all Tx antenna directions foreach Rx antenna direction. From Fig. I-2 (a), we can confirm an arrival wave (indicatedin red) with a path loss of 130 dB or less from directions other than approximately 230
degrees horizontal and approximately 15 degrees elevation, which are directions of theTx antenna as seen from the Rx antenna. Fig. I-2 (b) shows the minimum path loss inall Tx antenna directions for each Rx antenna direction at 160 GHz. Even at 160 GHz, it
is possible to confirm an arrival wave (shown in red) with a path loss of 120 dB or lessfrom the direction of approximately 230 degrees horizontal and approximately 15
degrees elevation corresponding to the Tx antenna direction and other directions fromthe Rx antenna. The arrival wave in the direction with an approximate 6 degree
elevation and approximate 40 degree horizontal is called A1, the arrival wave in the
direction with an approximate 6 degree elevation and approximate 110 degree horizontal
is called A2, the arrival wave in the direction with an approximate 24 degree elevation
and approximate 230 degree horizontal is called A3, and the arrival wave in the direction

11

with an approximate 6 degree elevation and approximate 350 degree horizontal is called
A4.
Similarly, Fig. I-3 (a) and Fig. I-3 (b) show the minimum path loss in all Tx antenna
directions for each Rx antenna direction at 160 GHz and 300 GHz at measurement point
B. As shown in Fig. I-3 (a), an arrival wave, which is yellow on the color bar, with a pathloss of approximately 120 dB can be seen from the angle of elevation of approximately 6
degrees and other directions at approximately 230 degrees horizontal, which is the
direction of the Tx antenna. As shown in Fig. I-3 (b), even at 300 GHz, a red arrival
wave with a path loss of 130 dB or less can be confirmed from the direction with an angleof elevation of approximately 6 degrees and other directions at approximately 230
degrees horizontal, which is the direction of the Tx antenna. The arrival wave in the
direction with an angle of elevation of approximately 12 degrees at approximately 50
degrees horizontal is referred to as B1, the arrival wave in the direction with an angle ofelevation of approximately 12 degrees at approximately 130 degrees horizontal isreferred to as B2, the arrival wave in the direction with an angle of elevation of
approximately 12 degrees at approximately 230 degrees horizontal is referred to as B3,
and the arrival wave in the direction with an angle of elevation of approximately 9

110

15

A1

A2

Tx

A4

150

0

-12

130

0

60

120
180
270 300
Angle of Azimuth (deg.)

360

170

30

100
A3

15

A1

A2

Tx

A4

140

0

-12

120

0

60

120
180
270 300
Angle of Azimuth (deg.)

360

Path Loss (dB)A3Angle of Elevation (deg.)

30

Path Loss (dB)Angle of Elevation (deg.)

degrees at approximately 320 degrees horizontal is referred to as B4.

160

B1

15

B2

B3

120

B4

Tx

140

0

-12

0

60

120
180
270 300
Angle of Azimuth (deg.)

360

160

Angle of Elevation (deg.)

100

30

110

B1

15

B2

B3

B4

130

Tx

150

0

-12

0

60

120
180
270 300
Angle of Azimuth (deg.)

360

170

(b) Composite diagram at 160 GHz(a) Composite diagram at 300 GHz
Fig. I-3. Measurement results at measurement point B.

12

Path Loss (dB)30

Path Loss (dB)Angle of Elevation (deg.)(a) Composite diagram at 300 GHz(b) Composite diagram at 160 GHz
Fig. I-2. Measurement results at measurement point A.

I-4. Conclusion
Using the two frequencies of 160 GHz and 300 GHz, which are sub-terahertz wave
bands, multipath wave measurements were carried out in an indoor environment. For
both frequencies, multiple arrival waves were confirmed from the wall surface andceiling directions except for the Tx antenna direction. The difference in the path loss of
the arrival wave from the wall and ceiling directions was approximately 10 dB toapproximately 30 dB compared to that for the arrival wave from the Tx antenna direction.
We found that even if the direct wave is blocked in the sub-terahertz wave bands of 160
GHz and 300 GHz, there is a possibility of communications using these arrival waves.
REFERENCE[1] NTT DOCOMO, INC., “White paper: 5G Evolution and 6G,” Jan. 2022.
https://www.docomo.ne.jp/english/binary/pdf/corporate/technology/whitepaper_6g/D
OCOMO_6G_White_PaperEN_v4.0.pdf[2] M. Inomata et al., “Terahertz propagation characteristics for 6G mobile
communication systems,” 2021 15th European Conference on Antennas andPropagation (EuCAP), Dusseldorf, Germany, pp. 1-5, March 2021.
[3] T. S. Rappaport et al., “Wireless communications and applications above 100 GHz:
Opportunities and challenges for 6G and beyond,” IEEE Access, vol. 7, pp. 7872978757, 2019.
[4] M. Nakamura et al., "Measurement of Multipath Waves at 160 GHz and 300 GHz inan Indoor Conference Room," 2023 17th European Conference on Antennas andPropagation (EuCAP), Florence, Italy, pp. 1-5. May 2023.
[5] "ITU-R M. 2412–0", Guidelines for evaluation of radio interface technologies for IMT2020, Oct. 2017.

13

II. Path Loss Characteristics from Microwave to Sub-THz Bands in Urban
Environment for Beyond 5G
Minoru Inomata, Wataru Yamada, Ryotaro Taniguchi, Daisei Uchida
NTT Access Network Service Systems Laboratories, NTT Corporation.
Nobuaki Kuno, Koshiro Kitao, Takahiro Tomie, Satoshi Suyama
NTT DOCOMO, INC.

Abstract—Beyond 5G, sub-terahertz bands, millimeter-wave bands, microwave bands,
path loss characteristics, frequency dependency
II-1. Introduction
New spectrum used for 6G and beyond 5G are sub-terahertz (THz) bands above 100
GHz for extremely high-speed communication data rates exceeding 100 Gbps. The ITUR Working Party 5D (WP5D) has agreed on a new recommendation on the Frameworkand overall objectives of the future development of IMT for 2030 and beyond and agreed
a report on the technical feasibility in bands above 100 GHz [1]-[4]. 5G have been rolled
out using a combination of sub 6 GHz and millimeter wave (mmW) band. And as
networks move toward beyond 5G, newly exploiting the mid-bands in 7-24 GHz and the
sub-THz bands, utilization of the combination of multiple frequency bands frommicrowave to sub-THz bands will be necessary for achieving extremely high-speed andreliable communications. In this study, aiming to exploit the new spectrums for beyond
5G, we reported the path loss, namely basic transmission loss, and building penetration
loss from microwave to sub-THz bands in urban environment [5]-[8].
II-2. Key Propagation Phenomena Affecting Radio Channel Characteristics in bands
above 100 GHz
Fig. II-1 shows the key propagation phenomena that affect channel characteristics forthe sub-THz bands in urban environment. ITU-R M.2412 [9] proposed models that cancalculate path loss that building shadowing and human blockage is considered and thebuilding penetration loss in accordance with the mixing ratio of building materials. The
path loss governs the coverage distance and the interference levels for the coexistence ofradio services using the same frequency bands. However, the M.2412 path loss models
are constructed by using measurement frequency bands below sub 6 GHz and mmW
bands [9]. Therefore, it is necessary to clarify the path loss characteristics, which
includes frequency dependency from microwave to sub-THz bands for operating the
combination of multiple frequency bands.

14

Attenuation byatmospheric gases

Rain attenuation
Base station

Vegetation
attenuation

Building penetration loss
Scatteringfrom roughsurface
Mobile station
Channel non-stationarity and loss due tostructure, vehicle and human blockage

Fig. II-1. Key propagation phenomena in bands above 100 GHz [5][6]II-3. Measurement System Setup
The measurement system uses frequency bands at 2, 26, 97, 158 and 300 GHz.
Continuous wave (CW) signal was transmitted using the omni-directional antenna [7][8].
The omni-directional antenna with 60° of half power beam width (HPBW) for 2, 26, 97
GHz in the elevation plane, 20° of HPBW for 158, 300 GHz. At the 2 GHz, the Rx omnidirectional antenna measured the path loss. At the 26, 97, 158 and 300 GHz, Rx
measured the CW signal while the cassegrain antenna with 4° of HPBW for 26 GHz, 2°of HPBW for 97 GHz, 1° of HPBW for 158 and 300 GHz was rotated 360 degrees in theazimuth plane and up to 20 degrees in the elevation plane. Path loss equivalent to theomni-directional antenna is obtained by this measurement method.
II-4. Path Loss Characteristics
The path loss measurement environment around Tokyo Station is shown in Fig. II-2
[7]. Path loss was measured at measurement points of the LOS and NLOS environment.
The Tx antenna was set on the rooftop of a measurement vehicle with height of 2.55 m,which is lower than the surrounding building’s rooftop level. The Rx antenna was fixed
to the trolley, with height of 1.7 m. We compared the free space path loss (FSPL)𝐹𝑆𝑃𝐿(𝑓, 𝑑) = 32.4 + 20 log10 𝑓 + 20 log10 𝑑 , M.2412, and measurement results. M.2412
path loss model was used a mandatory model for IMT-2020. The close-in (CI) model isalways used to characterize the path loss for LOS and the Alpha-Beta-Gamma (ABG)
model for NLOS. The CI model and ABG model are expressed by following equation:
𝑃𝐿𝐶𝐼 (𝑓, 𝑑)(𝑑𝐵) = 𝑃𝐿𝐹𝑆𝑃𝐿 (𝑓, 1) + 10𝑛 log10 𝑑 + 𝑋𝜎𝐶𝐼
where 𝑛 is the path loss exponent (PLE), and 𝑋𝜎𝐶𝐼 is a shadow fading with a standard
derivation of 𝜎 in dB.
𝑃𝐿𝐴𝐵𝐺 (𝑓, 𝑑)(𝑑𝐵) = 10𝛼 log10 𝑑 + 𝛽 + 10𝛾 𝑙𝑜𝑔10 (𝑓) + 𝑋𝜎𝐴𝐵𝐺
1 GHz

where 𝛼 and 𝛾 are the PLE and frequency dependence on path loss, respectively, 𝛽 isthe optimized offset in path loss. From Fig. II-3 (a) for LOS, the measured path loss
indicate the similar tendency as the predicted path loss of FSPL. It is obvious that the15

direct wave in the LOS environment was dominant. Therefore, the PLE for M.2412 pathloss model and PLCI is equivalent to that for FSPL. From Fig. II-3 (b) for NLOS, PLABG
by regressing the measured path loss had steeper PLE and larger frequency dependency
than the calculation results using M.2412 path loss model. In urban NLOS environment,
the diffracted and reflected waves with multiple reflection were major paths. Since thediffraction loss depends on the wavelength and the frequency dependency is significantly
larger than M.2412 path loss model. Specifically, in the case of scattering from a building
rough surface, the higher the frequency, the larger the scattering power and the lower
the reflection power, so it is assumed that since larger reflected loss occurred, the pathloss had steep distance attenuation.

Approx.
110 m

Tx omni-directional antenna
at 158-GHz band

Approx.
80 m
Approx.
90 m

Approx.
45 m
Approx.
Approx. 15 m
20 m

Rx

Tx

Rx cassegrain antenna androtator at 158-GHz band

Approx.
45 m

Approx. 82 m

Approx.
80 m

Approx. 34 m
Approx. 135 m

Approx. 150 m

Approx.
122 m

Approx.
70 m

Rx cassegrain antenna androtator at 300-GHz bandTx omni-directional antenna
at 300-GHz band(a) measurement system setup(b) measurement routes

Fig. II-2. measurement system setup and routes [7]Table II-1. Comparison results [7]
LOS

PLCI
FSPL
M.2412NLOS

PLABG
M.2412nConstant loss (dB)

𝑿𝑪𝑰𝝈 (dB)

1.9
2.0
2.1
α
4.34
3.53

32.4+20log10 f
32.4+20log10 f
32.4+20log10 f
β (dB)
γ
-4.1
2.52
22.4
2.13

6.04
6.21
6.86
𝑋𝜎𝐴𝐵𝐺 (dB)
6.19
9.22

16

Approx.
20 m

Measurement data at 2 GHz× Measurement data at 26 GHz

Measurement data at 158 GHz
Measurement data at 300 GHz

M.2412 at 2 GHz
M.2412 at 26 GHz
M.2412 at 158 GHz
M.2412 at 300 GHz

CI model for LOS and ABG model for NLOS at 2 GHz
CI model for LOS and ABG model for NLOS at 26 GHz
CI model for LOS and ABG model for NLOS at 158 GHz
CI model for LOS and ABG model for NLOS at 300 GHz

60

40

80

CI model80
FSPL

M.2412

100

Path loss (dB)Path loss (dB)

60

120
140

FSPL at 2 GHz
FSPL at 26 GHz
FSPL at 158 GHz
FSPL at 300 GHz

ABG model

100
M.2412

120
140

10

160

100
Tx-Rxdistance (m)(a) LOS40

50

60 70 80 90100
Tx-Rxdistance (m)200(b) NLOS

Fig. II-3. measurement results [7]II-5. Building Penetration Loss
This section introduces the results of measuring building penetration loss at 2, 26,
97, and 158 GHz in an urban environment in Tokyo [8]. Fig. II-4 shows the Tx andRx antenna setup. The Tx antenna was set on the roof of the building with a height of
2.24 m from the floor, and the Rx antenna was fixed to a trolley in rooms of the building
with a height of 1.5 m from the floor. Fig. II-4 shows a comparison between themeasurement building penetration loss, prediction results using the M.2412 low-loss
model, and proposed model. Table II-2 shows the components of those models. Typicalbuilding façades are composed of several materials. The measurement results in
M.2412 model can roughly be grouped into two categories: low building-penetration
loss results for old buildings with standard glass, and high ones for modern buildings
with infrared reflective glass. The proportion of measurement building composed ofmulti-layer standard glass was about 70%, and that of concrete was about 30%, and thematerial composite was similar to the low-loss model. Building penetration loss wascalculated by sum of the building penetration loss through the external wall PLtw
and the indoor loss dependent on the depth into the building PLin. PLtw consists ofan additional loss due to the non-perpendicular incidence and linear loss as afunction of frequency. When the propagation is deeper into the indoor room, indoor
loss PLin and variation due to the shadowing of the furniture can occur. It was foundthat measurement loss due to the non-perpendicular incidence and indoor loss was
equivalent to M.2412, but the measurement results were more gradually frequencydependent than M.2412. Since the material loss tends to increase as the frequency
becomes higher or permittivity becomes larger, the power of the transmission paths
through concrete is significantly lower in bands above 100 GHz. Thus, it was

17

assumed that the transmission paths through the glass were dominant and that theM.2412 low-loss model overestimated the building penetration loss due to the higher
transmission loss of the concrete. Therefore, we proposed an effective building
penetration loss model based on single transmission paths with the lowest
transmission loss through standard glass and the RMSE value was 5.7 dB improved.
Measurement data

Prediction results PLtw + PLin w/ prop.

× Prediction results PLtw + PLin w/ M.2412

Rx cassegrainantenna at 158 GHz

80
Building penetration loss (dB)Tx omni-directional
antenna at 158 GHz

70
60
50
40
30
20
10
0

(a) measurement system setup

Prediction results PLtw w/ prop.
Prediction results PLtw w/ M.2412

1

10Frequency (GHz)100(b) measurement results

Fig. II-4. measurement system setup and results [8]Table II-2. Comparison results [8]Path loss through external wall: PLtw (dB)Material loss:
Lglass (dB)

Indoor loss:
PLin (dB)

RMSE (dB)

M.24125 − 10 log10 (0.3 ∙ 10−𝐿𝑔𝑙𝑎𝑠𝑠⁄10 + 0.7 ∙ 10−𝐿𝑐𝑜𝑛𝑐𝑟𝑒𝑡𝑒⁄10 )2 + 0.2𝑓

0.5

11.3

Prop.

4.58 + 𝐿𝑔𝑙𝑎𝑠𝑠

2.58 + 0.15𝑓

0.28

5.7

II-6. Conclusion
Path loss and building penetration loss characteristics were investigated frommicrowave to sub-THz bands in urban environment. In the basic outdoor path loss, theresults indicate that since the direct wave in the LOS environment was dominant, thePLE for the CI model is equivalent to FSPL and M.2412. From results for NLOS, thePLE and frequency dependency were more significant than M.2412 due to larger
diffraction and reflection loss. It was found that the results of building penetration loss
above 100 GHz were more gradually frequency dependency than M.2412, and it was
assumed that the transmission paths through the glass were dominant. It isessential for accurate path loss prediction to clarify the dominant paths considering
the key propagation phenomena such as scattering from a building rough surface
and material loss in frequency bands above 100 GHz.

18

REFERENCE[1] NTT DOCOMO, INC. “White Paper, 5G Evolution and 6G,” Feb. 2021.
[2] ITU-R WP5D, “The ITU-R Framework for IMT-2030,” July.2023.
[3] ITU-R WP5D, “Preliminary draft new Report ITU-R M.[IMT.Above 100 GHz]
Technical feasibility of IMT in bands above 100 GHz,” July 2023.
[4] D. Uchida, Technologies and Proof-of-Concept Activities for 6G 2022 (TPoC6G 2022)
keynote speech, Finland, June 2022.
[5] M.Inomata, et al., “Sub-Terahertz Propagation Characteristics up to 150 GHz for 6G
Mobile Communication Systems,” EuMA International Journal of microwave andwireless technologies, March 2022.
[6] M. Inomata et al., "Terahertz Propagation Characteristics for 6G Mobile
Communication Systems," EuCAP2021, Germany, March 2021.
[7] M. Inomata et al., " Path-loss Characteristics and Modeling for 2-300-GHz bands in
Urban Microcell Environment," EuCAP2023, Italy, March 2023.
[8] M.Inomata et al., “Building Penetration Loss for 2 -150 GHz Bands for Pioneering
Terahertz Wave Bands of 6G,” IEICE Technical reports, vol. 122, AP-339, pp. 74-79,
January 2023 (Japanese edition).
[9] Rep. ITU-R M.2412-0, “Guidelines for evaluation of radio interface technologies for
IMT-2020,” Oct. 2017.

19

III. Evaluation Technique for Propagation Characteristics of 300-GHz-band
Radio wave near the Human Body
Akihiko Hirata, Chiba Institute of Technology

Abstract—In order to put a terahertz-band Body Area Network (BAN) into practicaluse, it is necessary to measure a large amount of 300 GHz-band radio propagation data,such as the direction of arrival (DOA) of 300 GHz-band radio signals near the human
body, and to construct a propagation model. This chapter describes the evaluation
techniques for 300-GHz radio propagation near the human body, including diffraction
and reflection characteristics near the human body.
III-1. Introduction
In Beyond 5G, the next generation of mobile communications, the use of 300 GHz band
wireless links, which can achieve data rates of over 100 Gbit/s, is currently being
considered for various use cases such as mobile access, wireless local area networks(LANs), and body area networks (BANs). For practical application of these use cases, itis necessary to evaluate the radio propagation characteristics near the human body inthe 300 GHz band and to construct a radio propagation model at this frequency. We havebeen conducting radio propagation experiments and simulations for building radio
propagation models for each of the above-mentioned use cases. In this paper, we report
the results of these experiments and simulations of radio propagation at 300 GHz.
III-2. Shadowing loss due to human body
A 300-GHz-band transmitter (Tx) and a receiver (Rx) were used to measure theshadowing loss due to the human body [1]. The unmodulated continuous wave with afrequency of 293.4 GHz was used for the evaluation of propagation characteristics nearhuman body. The Tx output power was 1.2 mW. A waveguide probe antenna with a gainof 7 dBi was used for both Tx and Rx. The distance between the Tx and Rx was fixed at
2 m, and the received power fluctuation was measured when the human body was moved
horizontally from -35 cm to +35 cm in 5 cm increments at the midpoint between thetransmitter and receiver. Fig. III-1(a) shows the offset position dependence of the
shadowing loss due to the human body. When the human body was facing the Tx, theshadowing loss at an offset position of 0 cm was about 26 dB.
To verify the validity of the measurement results, a radio propagation simulation using
the ray trace method was performed. The results are also shown in Fig. III-1(a). Whena human model (Model A) with closed armpits shown in Fig. III-1(b) was used, adeviation of 18 dB from the experimental results was observed at an offset position of 0
cm. Fig. III-1(a) also shows the simulation results using a human body model with a 1

20

cm gap between the armpits and the torso (Model B); for Model B, the simulated
shadowing loss is close to the experimental results when the offset position is 0 cm. Whenthe offset position is at the gap between the arm and the torso, the simulated shadowing
loss is less than 10 dB, which is 15 dB less than the experimental result. This may be
because the position of the gap between the arm and the torso is different between thesimulation and the experiment.

Fig. III-1. (a) Dependence of the shadowing loss by human body on the offset position,(b) human model with armpit closed, (c) human model with a 1 cm gap between the armand the torso.
III-3. 300 GHz band human body phantom
To construct a radio propagation model near the human body, it is necessary tomeasure the shadowing, reflection and scattering characteristics of radio propagationnear human body. Since it is difficult for a living human to maintain the same position
for a long period of time, it is necessary to use a human body phantom with the same
complex permittivity as that of the living human body at the measurement frequency inorder to obtain reproducible data. However, there has been no 300-GHz-band humanphantom with the same complex permittivity as that of the living human body. Yamazaki
et al. reported that the real and imaginary dielectric constants of the human body in the
300 GHz band will be 3.5 to 5.0 and 2.0 to 3.0, respectively, in 2022 [2]. We fabricated a
new material based on silicon rubber and evaluated its complex permittivity byellipsometry using the THz time-domain spectroscopy (TDS) technique. Fig. III-2(a)shows the complex permittivity of the prototype human phantom at 200 to 400 GHz. The
real and imaginary dielectric constants are 4.6 and 1.2, respectively, which are close to
those of a living human. We prototyped a human phantom with the shape of a humanbody based on this silicon rubber [3]. Fig. III-2(b) shows a photograph of the prototype
human phantom. In the following sections, the diffraction and reflection characteristics
at 300 GHz near the human body are evaluated using this human phantom.

21

Fig. III-2. (a) Complex permittivity measurement results of the prototype humanphantom, (b) Photograph of the prototype human phantom.
III-4. DOA measurement result
Terahertz-band DOA measurements require large devices that change the azimuthand elevation angles of the Rx. However, these devices take time to measure and are
difficult to place near the human body. We have developed a high-speed terahertz
hemispheric scanner capable of making DOA measurements of radio propagation at 300
GHz [3]. Fig. III-3(a) shows a schematic diagram of the terahertz band hemisphericscanner. The scanner consists of a 300 GHz-band Rx, a rotating mirror, a fixed mirror,
and a rotating stage. 35 dBi lens antenna is connected to the Rx. The rotating mirror
and the rotating stage control the elevation and azimuth angles of the terahertz beam tobe measured, respectively. The scanning range of the azimuth angle is 360o, and thescanning range of the elevation angle can be selected from 0 to 90° or 45 to 135° bychanging the adapter. Fig. III-3(b) shows a photograph of the terahertz hemisphericscanner. The outer surface of the scan direction is surrounded by a radio wave absorber
to avoid stray light reception outside the scan direction.
The diffraction characteristics of 300 GHz band radio propagation near the humanbody were evaluated using a terahertz band hemispherical scanner and a humanphantom. The Tx and Rx were placed opposite each other at a transmission distance of
1 m, and a human phantom was placed at the midpoint between the transmitter andreceiver. The human phantom was placed directly opposite to the transmitter. A 7 dBi
waveguide probe antenna was used for the Tx. Fig. III-4(a) shows the results of DOA
measurements. Two diffracted propagation paths were observed on the outside of botharms. In addition, two propagation paths that pass through the gap between the armand the torso were observed. The DOAs of the two propagation paths through the gap
between the arm and torso were wider than the DOAs of the propagation paths throughthe outer arms.
Next, Fig. III-4(b) shows the results of DOA measurements when the human phantom
was wearing a down coat. When the human phantom wears the down coat, these paths

22have wider DOA perpendicular to the ground than when the human phantom does notwear the down coat. In addition, periodic variations in DOA were observed, reflecting
the periodic surface geometry of the down coat. These wider DOAs may be due to thescattering of radio waves inside the down coat.

Fig. III-3. (a) Schematic diagram of the hemispheric scanner, (b) Photograph of thehemispheric scanner.

Fig. III-4. (a) Photographs of the experimental setup and measurement results of DOA
due to the diffraction on the human phantom, (b) Photographs of the experimental
setup and measurement results of DOA due to the diffraction on the human phantom
wearing a down coat
Next, the reflection characteristics on the human body were evaluated using thehemispherical scanner and a human body phantom. The Tx and Rx were placed at 0.5 m
from the human body phantom so that the angle of incidence and the angle of reflection
were 45o. Fig. III-5(a) and Fig. III-5(b) show the measurement results of DOA of thereflected wave from the human body phantom without and with a down coat, respectively.
In the case of the phantom without a down coat, the DOA of the propagation path of thereflected wave was observed separately for each body part, such as the chest, chin, andarms. When the down coat was worn, the DOA was observed over a wider area and23

reflections were no longer observed separately for each body part. This may be due to thescattering of the down coat.

Fig. III-5. (a) Photographs of the experimental setup and DOA measurement results
during reflection from the human phantom, (b) Photographs of the experimental
system and DOA measurement results during reflection from the human phantom
wearing a down coat.
III-5. Conclusion
Evaluation of radio propagation characteristics near the human body was conducted.
To obtain a large amount of reproducible radio propagation data, a human body phantom
with a complex permittivity close to that of the human body at 300 GHz was developed.
The reflection and diffraction characteristics of radio wave propagation near the humanbody were evaluated, and it was found that the gap between the arm and torso and theclothing had a significant effect on the radio propagation characteristics near the humanbody.
III-6. Acknowledgements
These research results were obtained from commissioned research (No.00401) by the
National Institute of Information and Communications Technology (NICT), Japan.
REFERENCE[1] S. Takagi et al., “Shadowing and Reflection Loss due to Human Body in 300-GHzband Wireless Links,” IEICE, B-1-13, 2022.
[2] S. Yamazaki et al., “Investigation of Biological Phantom Reproducing Dielectric
Properties of Skin Tissue in Terahertz Frequency Range,” IEICE Technical Report
PEM, p1, 2022.
[3] A. Hirata, "Measurement of 300 GHz Band Radio Propagation Characteristics in theVicinity of Human Body using Terahertz Hemispheric Scanner," 2023 IEEE APS and
USNC-URSI, pp. 1221-1222, 2023.

24

IV. AI Calibrationtransmission

Techniquefor

RF

Impairments

with

Sub-THz

Keita Kuriyama, Kentaro Tanaka, Hitoshi Hasegawa, Toshifumi Miyagi
NTT Access Network Service Systems Laboratories, NTT Corporation
Satoshi Suyama, Huiling Jiang, Atsuya Nakamura
6G Network Innovation Department, NTT DOCOMO, INC.

Abstract— In the sixth-generation mobile communication system (6G) era, highfrequency bands (e.g., sub-terahertz (sub-THz) bands) show promise for achieving
extremely high-speed and high-capacity communications. However, it is difficult toensure radio frequency (RF) circuit quality in high-frequency bands compared to lower
frequency bands below mmWave, and utilizing higher performance and higher quality
circuits lead to higher costs. Considering the increasing popularity of the high-frequency
bands, it is vital to achieve low cost while simultaneously ensuring communication
quality. It is also necessary to optimize the communication quality, cost, and power
consumption of the whole radio access network (RAN) by considering the diversified
topology. This chapter presents a conceptual overview of our proposed artificial
intelligence (AI) device calibration and the AI calibration network to optimize
communication quality and calibration cost based on AI. Further, as a basic examinationof AI device calibration, we demonstrate a compensation technology for RF impairments
based on a deep neural network (DNN).
IV-1. Introduction
One of the requirements for sixth-generation mobile communication systems (6G) isan extremely high data rate and capacity. Radio access technologies (RAT) to provide
data rates over 100 Gbps are being discussed as a potential way of meeting thisrequirement. To achieve 100 Gbps, exploiting higher frequency bands between 100 GHz
and 300 GHz with a wider bandwidth than 5G (e.g., sub-terahertz (sub-THz) bands) is apromising approach. However, to utilize the sub-THz bands in 6G, similar to the casewhen introducing the millimeter-wave band in 5G, there are many technical issues thatneed to be resolved by the 2030s. These technical issues are diverse and exist mainly infour areas: radio propagation, radio frequency (RF) devices, modulation anddemodulation schemes, and air interfaces, which are summarized in detail for each areain [1].
Focusing on the technical issues in RF devices, the characteristics of the RF devices
mainly depend on the frequency band and the signal bandwidth, and it is difficult toensure the same circuit characteristics in the high-frequency bands as those in the low
frequency [2]. Some RF impairments, such as frequency selectivity, IQ imbalance, direct25

current (DC) offset, carrier leakage, phase noise, and nonlinear distortion, have become
increasingly pressing as implementation challenges because they can cause degradationof the communications quality. While performance enhancement and high integration of
the RF devices are required, for the popularization of high-frequency bands, the RF
devices must be manufactured with a level of accuracy and a cost that enables usage in
6G commercial services. Cost reduction can be achieved by allowing the use of lowquality devices, but a calibration scheme by digital signal processing (DSP) is necessaryfor ensuring the communication quality. Although a partial compensation technique for
the RF impairments by DSP has been proposed, this technique requires designing an
optimal digital calibration for high-frequency bands, where the influence of each RF
impairment is both large and mixed. In recent years, technology that compensates for
multiple RF impairments by utilizing artificial intelligence (AI) such as deep neuralnetworks (DNN) has attracted attention [3].
The performance of the RF device and the resources (e.g., processor capability andpower consumption) available for DSP are different for each wireless device. Therefore,
it is desirable that the cost of digital calibration be dynamically optimized in accordancewith the constraints of the available hardware and the required quality from theapplications. In addition, it is necessary to optimize the communication quality, cost, andpower consumption of the whole radio access network (RAN) by considering thediversified topology. In this chapter, we present our concept of AI device calibration and
the AI calibration network [4], which utilizes AI to optimize the communication quality
and the calibration cost. Also, as a basic technology of AI device calibration, ademodulation technology [5] utilizing DNN is introduced.
IV-2. Concept of AI calibration
IV-2.1. AI device calibration
Fig. IV-1 shows the concept of AI device calibration. The required specifications forthe equipment differ between base station (BS) and user equipment (UE), which means
the characteristics of the RF circuit, the capability of the processor, and the allowable
power consumption for DSP are also different from each device. On the other hand,regarding the communication quality, the total communication quality that can beobserved end-to-end only needs to meet the requirements. In other words, points such as
calibrating within the transmitting station to send a high-quality signal, calibrating onlyat the receiving station, setting the proportion of each calibration processing load, etc.
can be freely designed. In AI device calibration, the existence of digital calibration ineach station, the compensation scheme to be used, and the level of compensation
accuracy are dynamically controlled in accordance with the constraint of available

26

hardware and the demands of the communication quality. As a case study of acompensation scheme, we introduce a DNN demodulator in Section IV-3.
(Transmitter)

Modulation
circuit(Receiver)

Processor

Processor

Calibration
circuitCalibration
circuitRF circuit

RF circuit

Demodulation
circuit

Optimization

Hardware andresource info.

AI device calibration

Fig. IV-1. Concept of AI device calibration.
IV-2.2. AI calibration network
In the 6G era, many devices will be connected to the RAN through various frequency
bands, so the topology of the RAN is expected to diversify. Therefore, new radio network
topology (NRNT) has been investigated to improve the performance of RAN for 5G
Evolution and 6G [6]. In NRNT, the topology changes dynamically in accordance withthe environment, situation, and requirements based on various key performanceindicators (KPIs). With sub-THz bands, optimization in the whole of RAN including therelay station (RS) is required, since the impact on KPIs such as equipment cost andpower consumption seems to be large.
Fig. IV-2 shows the concept of the AI calibration network. The AI device calibration
described in Section IV-2.1 is extended to the entire RAN. The AI calibration network
collects the resource information such as the performance of the RF devices of eachequipment, processor capability, and power consumption available for DSP. The
calibration cost of each device is controlled in accordance with the constraint of available
hardware and the requirement of communication quality. In addition, appropriate routeselection is performed for the optimization of the whole network.
RS
RS

BS
UE

BS

RS

Hardware, resource,and environment info.

UE

Optimization

AI calibration network

Fig. IV-2. Concept of AI calibration network.

27

IV-3. Compensation technique based on AI
IV-3.1. Deep neural network demodulator
In this section, we introduce a technique to compensate for multiple RF impairments
based on DNN. Fig. IV-3 shows the system model of our DNN demodulator. Assuming
single carrier-(SC) frequency domain equalization (FDE) transmission with nonlinear
distortion of the amplifier and IQ imbalance of the IQ modulator/demodulator, bit
detection by DNN is performed for the received symbols after FDE. The FDE output isdivided into an IQ real valued sequence after IFFT processing and passed to the input
layer of the DNN. The DNN outputs a vector consisting of values from 0 to 1 via thesigmoid function. The DNN is trained to minimize the root mean square error (RMSE)of the training data and the output data. Round processing is performed after the DNN,

PA

Fadingchannel(Receiver)
IQ
demod.
IQ
ADC

LO

DNN Detect
bits
demod.

Round into
binary bits

IQ
DAC

IQ
mod.

FFT
…
FDE
…
IFFT

Input
bits

Generate
SC waveform(Transmitter)

Activation(sigmoid)and the bit string corresponding to the index of the input data is output.

LO

Real-valued data
after FDE

Q-ch
I-ch

Output2D conv.

Batch norm.

2D conv.

Input

Ｚ

Activation

１

１x

DNN

Fig. IV-3. System model of DNN demodulator.
IV-3.2. Numerical Results
In this section, we evaluate the effectiveness of the DNN demodulator. Table IV-1 lists
the parameters utilized during learning and validation. The nonlinearity of the power
amplifier utilizes a Rapp model with the input back off (IBO) of 2 dB. The IQ imbalanceof the quadrature modulator and demodulator is 1 dB in amplitude and 5° in phase. The
average signal to noise power ratio (SNR) is 30 dB while learning the DNN. The fading
channel assumes a static environment and utilizes common values in the learning andtest data. We use 16-QAM and 64-QAM for the modulation scheme.

28Fig. IV-4 shows the bit error rate (BER) performance of the DNN demodulator, w/ocompensation, and w/o RF impairments of nonlinear distortion and IQ imbalance. In w/ocompensation, hard decision demodulation is performed after IFFT. Compared to w/ocompensation, the DNN demodulator improves the BER performance and approaches
the characteristics of the case without RF impairments. These results demonstrate thatthe communication quality can be improved by using the DNN demodulator for RF
impairments.
Table IV-1. Simulation parameters for training and validation.
Parameter
Value
Modulation scheme
16-QAM, 64-QAM
PA Nonlinearity
Rapp model[7], p=2,
IBO=2 dBTx and Rx IQ imbalance
Amplitude: 1 dB
Phase: 5°
Channel model
Static, exponential
CIR length
4
Average SNR
Training: 30 dB
Validation: 10 dB – 30 dB
FFT size
64

BER

64-QAM

16-QAMw/o impairments

w/o compensation
DNN demod.
Average SNR [dB]Fig. IV-4. BER performance with PA nonlinearity, IQ imbalance, and fading channel.
IV-4. Conclusion
To popularize the use of high-frequency bands, it is desirable to optimize the trade-off
between communication quality and cost reduction in accordance with the application29

requirements. We introduced a conceptual overview of AI device calibration as an
optimization technology that dynamically controls the calibration cost in accordancewith the constraint of available hardware and the demand of the communication quality.
We also proposed an AI calibration network to optimize the device calibration cost acrossthe RAN, which is an important KPI in the sub-THz bands. Our findings demonstrated
the effectiveness of the RF impairment compensation technology based on DNN as abasic technology for AI device calibration.
REFERENCE[1] H. Fukuzono, S. Suyama, D. Lee, D. Uchida, T. Okuyama, M. Iwabuchi, J. Mashino,
and Y. Kishiyama, “Issues of Sub-Terahertz-Band Radio Access Technologies for 6G,”

IEICE Tech. Rep., vol. 121, no. 302, RCS2021-180, pp. 30–34, Dec. 2021.
[2] U. Gustavsson et al., “Implementation Challenges and Opportunities in Beyond-5G
and 6G Communication,” IEEE Journal of Microwaves, vol. 1, no. 1, pp. 86–100, Jan.
2021. DOI: 10.1109/JMW.2020.3034648[3] J. Pihlajasalo, D. Korpi, T. Riihonen, J. Talvitie, M. A. Uusitalo, and M. Valkama,“Detection of Impaired OFDM Waveforms Using Deep Learning Receiver,” 2022
IEEE 23rd International Workshop on Signal Processing Advances in WirelessCommunication(SPAWC),

Oulu,

Finland,

2022,

pp.

1–5.

DOI:

10.1109/SPAWC51304.2022.9834021[4] K. Tanaka, K. Kuriyama, H. Hasegawa, T. Miyagi, S. Suyama, and Takayuki Yamada,“A study of calibration techniques for the use of high frequency bands,” in Proc. IEICE
Society Conf., B-5-46, Sept. 2023.
[5] K. Kuriyama, K. Tanaka, H. Hasegawa, T. Miyagi, S. Suyama, and Takayuki Yamada,“A Study of Compensation for RF Impairments Based on DNN in SC-FDE
Transmission with Higher Frequency Bands,” in Proc. IEICE Society Conf., B-5-46,
Sept. 2023.
[6] M. Iwabuchi, S. Suyama, T. Arai, M. Nakamura, K. Goto, R. Ohmiya, D. Uchida, T.
Yamada, and T. Ogawa, “Concept and Issues of New Radio Network Topology for 5G
Evolution & 6G,” IEICE Tech. Rep., vol. 122, no. 235, RCS2022-148, pp. 101–106, Oct.
2022.
[7] T. Schenk, RF imperfections in high-rate wireless, Springer

30

V. Sub-THz-Band Massive MIMO Technology for Beyond 5G/6G
Satoshi Suyama
NTT DOCOMO, INC.

Abstract— A massive multiple-input multiple-output (M-MIMO) technology has been
introduced into 5G in millimeter-wave bands such as 28 GHz. For Beyond 5G/6G,
extremely high data rate exceeding 100 Gbps is required, and hence so-called subterahertz (sub-THz) band targeting the frequency from 100 GHz to 300 GHz has
attracted much attention. This paper introduces link-level simulations of a 100 GHzband radio access technology exploiting M-MIMO, over-100 Gbps downlink throughput
and coverage performances are clarified in the sub-THz band.
V-1. Introduction
Higher frequency bands such as 28 GHz millimeter wave (mmW) have been introduced
into 5G. It is also well known that beamforming (BF) technologies using massive
multiple-input multiple-output (M-MIMO) can compensate for larger pathloss in thehigher frequency bands. Towards Beyond 5G/6G, extremely high data rate exceeding 100
Gbps is expected to realize by exploiting much wider bandwidth of so-called subterahertz (sub-THz) wave targeting the frequency from 100 GHz to 300 GHz [1]-[3]. For
realizing 5G and 6G, a variety of studies on advanced M-MIMO technologies are being
carried out all over the world. NTT DOCOMO is also promoting to perform experimental
trials and computer simulations for them. This paper briefly introduces studies on the
M-MIMO technology for 6G. Concretely, it shows over-100 Gbps downlink (DL)throughput and coverage performances by 100 GHz-band link-level simulations
employing the M-MIMO technology.
V-2. M-MIMO Technology Based on Hybrid BF
This paper employs the M-MIMO technology based on hybrid BF scheme as shown in
Fig. V-1. The scheme is joint processing of analog BF and channel state information(CSI)-based digital precoding [3], [4]. Let denote NT and L the number of antenna
elements in M-MIMO and the number of analog beams in the hybrid BF, respectively. L
is same as the number of up-converter (down-converter) or digital processing chains. L
is set to 16, for example, in the case of NT = 4096, a subarray consisting of 256 (= NT/L)
antenna elements generates one beam. First, the hybrid BF selects each direction of L
analog beams based on the maximum received power criterion, and then the digital
precoding is performed for M-MIMO streams by exploiting the estimated CSI [3].

31NT /L
L

+CP

・・・

DAC

Upconverter

・・・

IFFT

DAC
・・・

+CP

・・・

・・・

#M

IFFT

#1

Upconverter

・・・

#1

Digital precoding

L

# NT

Analog BF

Fig. V-1. Configuration of M-MIMO based on hybrid BF.

V-3. Performance Evaluation of 100 GHz-Band M-MIMO by Link-Level Simulations
Link-level simulations evaluate performances of sub-terahertz 100 GHz-band M-MIMO
for 6G [5]. Table V-1 shows simulation specifications in the 6G link-level simulations.
The performances of 5G are also evaluated in addition to 6G. The center frequency fc
was set to 28 GHz and 100 GHz for 5G and 6G, respectively. A total bandwidth B = 400
MHz for 5G with 4 component carriers (CCs) and B = 6.4 GHz for 6G with 2 CCs, whichis 16 times of 5G. In order to extract the full performance of MIMO transmission by thedigital precoding, this paper employed OFDM as 5G and 6G radio access, and time slots
in time division duplex (TDD) were all assigned to DL, in other words, the ratio of timeslots in DL and uplink (UL) (TDD ratio) was 10 to 0. In 6G, the number of BS antenna
elements in M-MIMO, NT is set to 256, 1024, and 4096, and the number of MS antenna
elements, NR is made to be a planar array equivalent to 256 elements in which 16
uniform planar arrays with 16 elements and half-wavelength interval are arranged at
3.5-wavelength interval so as to have the same aperture size as the 5G MS antenna with
16 elements and one-wavelength interval. Note that the simulations in 6G were carriedout by 16-element uniform planar array with 3.5-wavelength interval, and 12 dB was
considered as an additional BF gain of MS in evaluating coverage performances. L is set
to 16, for example, in the case of NT = 4096, the subarray consists of 16 elements in thehorizontal direction and 16 elements in the vertical one. M is set to 16 for 5G and 4 and
8 for 6G. This simulation employed adaptive modulation and coding (AMC) using the set
with the highest throughput according to the average signal-to-noise power ratio (SNR)of each stream. When the transmission efficiency considering pilot insertion loss, etc. is
0.8, the maximum throughput becomes 28.96 Gbps in 5G, 115.85 Gbps and 231.70 Gbps
in 6G with M = 4 and M = 8, respectively. Since the UL throughput is expected to become
important in 6G, M = 8, which can allocate the TDD ratio to the UL to some extent, is amore realistic setting.

32

Table V-1.

SIMULATION SPECIFICATIONS
5G

6G

Center frequency, fc

28 GHz

100 GHz

Bandwidth, B

400 MHz

6.4 GHzNo. of CCs

4

2

60 kHz1920 kHzSubcarrier spacing

1584

No. of subcarriers
No. of FFT points

2048

No. of BS antenna elements, NT

256

256, 1024, 4096

No. of MS antenna elements, NR

16

Comparable to 256

No. of beams, L

16

No. of MSs, NU

1

No. of streams, M
Average angle(azimuth, zenith)
Angular spread(azimuth, zenith)
TDD ratio

16

4, 8

Departure: (90 deg., 90 deg.)
Arrival: (90 deg., 90 deg.)
Departure: (5 deg., 5 deg.)
Arrival: (20 deg., 20 deg.)
DL : UL = 10 : 0

Transmission efficiency
Modulation scheme
Channel coding

0.8
QPSK, 16QAM, 64QAM, 256QAM
Turbo code (coding rate: 1/2, 2/3, 3/4)

Transmission power, PT

30 dBmNoise power density, NPD-174 dBm/Hz

Noise figure, Nf
Multipath fading channel

5 dB
16-path Nakagami-Rice fadingwith Rice factor of 10 dB

V-4. Simulation Results of 100 GHz-Band M-MIMO Technology
V-4.1. Throughput Performances with Average SNR
Fig. V-2 shows DL throughput performances in 5G and 6G for the average received
SNR at the MS antenna. From this figure, the average SNR that can achieve 100 Gbps
in 6 G with NT = 256, 1024, and 4096 is 8.5 dB, 2.5 dB, and -0.5 dB, respectively. As NT
increases, the BF gain improves, however, in the case of NT = 4096, the BF gain decreases
with the expansion of antenna aperture size. The spectrum efficiency in 6G with M = 8
is half of that in 5G, while even in case of NT = 256, the effect of increasing the bandwidth
B by 16 is large, which shows that higher throughput can be achieved compared with 5G.
From these evaluation results, it was shown that the extreme wideband 6G can provide
extremely higher data rate compared with 5G [5].

33

240

L = 16, NU = 1

220

NT = 4096

User throughput (Gbps)

200
180

M=8

NT = 1024

160

140

NT = 256120

6G (fc = 100 GHz,
B = 6.4 GHz)100

NT = 4096, M = 480
60

5G (fc = 28 GHz,
M = 16, NT = 256)

40
20
0
-20

-15

-10

-5

0

5

10

15

20

25

30

Average SNR (dB)

Fig. V-2. Throughput versus average SNR.
V-4.2. Coverage Performances
By calculating the received SNR based on the link budget, the coverage performancesof 6G radio access using the center frequency of 100 GHz are evaluated [5]. When the
total transmission power is PT (dBm), the pathloss is LP (dB), the noise power density is
NPD (dBm/Hz), and the noise figure is Nf (dB), the received SNR  (dB) is expressed asΓ = 𝑃T + 𝐺BS + 𝐺MS − 𝐿P − (𝑁PD + 10 log10 𝐵 + 𝑁f ) ,where GBS and GMS denote antenna (element) gains of BS and MS, respectively. Here, inconsideration of the fact that the rectilinearity of 100 GHz is higher than that of 28 GHz,
the pathloss LP in the LOS environment of a rural area is used [6]. In this evaluation, it
was assumed that GBS = GMS = 3 dBi, and 6G considered 12 dB as the additional BF gainof MS.
Fig. V-3 shows the throughput for the distance between BS and MS as coverage
performances. As a reference, the coverage performance of 28 GHz-band 5G is also shown.
From this figure, the throughput of 100 GHz-band 6G decreases as the distance increases,
and 6G can achieve the throughput of 100 Gbps up to the distance of 75 m for M = 4 and120 m for M = 8. The results of M = 4 and M = 8 become closer as the distance increases.
This is because the MIMO spatial multiplexing is performed under the condition thatthe total transmission power is constant, so that the transmission power per stream of
M = 4 is twice that of M = 8. In comparison with 5G, 6G can achieve the throughput ofapproximately 118 Gbps even in the distance of 100 m by exploiting the extreme
wideband of 6.4 GHz despite the larger pathloss, and it is proven that almost 13 times
higher throughput than 5G can be achieved.

34

240
L = 16, NU = 1

220
User throughput (Gbps)

200
M=8

180
160
140

M=4

6G (fc = 100 GHz,
NT = 4096)

120
100
80
60

5G (fc = 28 GHz,
M = 16, NT = 256)

40
20
0

0

20

40

60

80

100 120 140 160 180 200

Distance (m)Fig. V-3. Coverage performance.
V-5. Conclusion
For Beyond 5G/6G, this paper introduced NTT DOCOMO’s recent studies on the subTHz-band M-MIMO technologies. Link-level simulations using 100 GHz-band M-MIMO
clarified that 6G can achieve extremely high DL user throughput over 100 Gbps. In
addition, 100 GHz-band system-level performances have been shown by NTT
DOCOMO’s developed 6G simulator employing multiple base station antennas with MMIMO in a future shopping mall environment [7].
REFERENCE
[1]

Press

releasefromNTT

DOCOMO

on

Jan.

24,

2020.

https://www.nttdocomo.co.jp/english/info/media_center/pr/2020/0124_00.html[2] NTT DOCOMO, INC., “White paper: 5G Evolution and 6G (version 5.0),” Jan. 2023.
[Online].

Available:

https://www.nttdocomo.co.jp/english/binary/pdf/corporate/technology/whitepaper_6g/
DOCOMO_6G_White_PaperEN_v5.0.pdf[3] S. Suyama, T. Okuyama, Y. Kishiyama, S. Nagata, and T. Asai, “A study on extreme
wideband 6G radio access technologies for achieving 100 Gbps data rate in higher
frequency bands,” IEICE Trans. Commun., vol. E104-B, no. 9, pp. 992-999, Sept. 2021.
[4] T. Obara, S. Suyama, J. Shen, and Y. Okumura, “Joint processing of analog fixed
beamforming and CSI-based precoding for super high bit rate massive MIMO
transmission using higher frequency bands,” IEICE Trans. Commun., vol. E98-B, no.
8, pp. 1474-1481, Aug. 2015.

35

[5] S. Suyama, T. Okuyama, N. Nonaka, and T. Asai, “Recent studies on massive MIMO
technologies,” IEEE RWW2022, Jan. 2022.
[6] 3GPP, TR 38.901 (V16.1.0), “Study on channel model for frequencies from 0.5 to 100
GHz (Release 16),” Dec. 2019.
[7] T. Okuyama, S. Suyama, N. Nonaka, and T. Asai, “6G system-level simulator -Toward
100 GHz band, 100 Gbps extreme-high-data-rate communications-,” NTT DOCOMO
Tech. Journal, vol. 23, no. 3, pp. 13–26, Jan. 2022.

36

VI. High-Capacity Wireless Transmission Technology using Orbital
Angular Momentum Multiplexing in the Sub-THz Frequency Bands
Doohwan Lee, NTT Network Innovation Laboratories, NTT Corporation
Hirofumi Sasaki, NTT Network Innovation Laboratories, NTT Corporation
Yasunori Yagi, NTT Network Innovation Laboratories, NTT Corporation
Riichi Kudo, NTT Network Innovation Laboratories, NTT Corporation

Abstract— Experimental demonstrations of more than 1-Tb/s wireless transmissions
based on orbital angular momentum (OAM) multiplexing using Butler matrices andparabolic reflector antennas were conducted in sub-terahertz (sub-THz) frequency bands.
We designed and implemented an 8×8 Butler matrix that performs discrete Fouriertransform (DFT) processing in an analog way to generate and separate OAM waves withhigh precision over a 32-GHz bandwidth. Also, a parabolic reflector antenna wasdesigned and implemented to extend the transmission distance. An experiment was
conducted in a shielded room, and 1.4 Tb/s wireless transmissions were achieved usingeight OAM modes and dual polarization (16 signals) at distances of 1 m and 20 m—without and with parabolic reflector antennas, respectively.
VI-1. Introduction
As the data traffic of wireless communications continues to grow exponentially, highcapacity wireless transmissions are in great demand for many scenarios. Several
generations of wireless networks have emerged and been passed on, and discussions
related to the sixth generation (6G) are currently ongoing. The aggregated wirelesstransmission data rate in these networks has increased up to 1,000 times as thegenerations have migrated over the last decade, and over 200 Gb/s is expected in thenext generation of wireless networks [1]. The capacity of backhaul and fronthaul
connections for these networks has to be enlarged accordingly, and providing suchconnections wirelessly is useful in terms of the maintenance and deployment flexibilityof base stations. To achieve further high-capacity wireless transmissions, we need toutilize sub-terahertz (sub-THz) bands in 6G, where a wide bandwidth is available. New
technology such as orbital angular momentum (OAM) multiplexing is emerging toprovide high-capacity wireless transmissions utilizing a wide bandwidth in sub-THz
frequency bands.
VI-2. OAM Multiplexing Transmission and Use Cases
OAM is one of the physical properties of electromagnetic waves characterized by aspiral phase front in the direction of propagation. The phase rotation speed of the helicalphase front is called the OAM mode. The transmission capacity can be enhanced with37

different data signals using different OAM modes. Fig.

VI-1 (a) illustrates OAM

multiplexing transmissions using three different OAM modes. This technology should
enable high-capacity wireless transmissions comparable to those of optical transmission
systems envisioning various applications, including wireless backhaul/fronthaul
between base stations, as shown in Fig. VI-1 (b). The eventual development of thistechnology will likely support the creation and spread of various future services, such as
virtual reality/augmented reality (VR/AR), high-definition video transmissions,connected cars, and telemedicine, as innovative wireless-communication technology inthe 6G era.

(a)

(b)

Fig. VI-1. (a) Illustration of OAM multiplexing transmissions, (b) Use case:complement of optical fiber connections.
VI-3. Enabler of OAM Multiplexing in Sub-THz Frequency Bands
VI-3.1. Butler Matrix Design for Wideband Transmissions
Electromagnetic waves with different OAM modes can be generated and separated
simultaneously using a uniform circular array (UCA) with discrete Fourier transform(DFT) processing. Such processing can be performed using an analog approach [2, 3] ora digital one [4]. We chose the analog approach to reduce the digital signal processing
burden and to utilize the wide bandwidth of sub-THz frequency bands fully.
Fig. VI-2 (a) shows a schematic of our designed 8×8 Butler matrix that performs DFT
processing in an analog way [2]. It is composed of wideband 3-dB couplers (π/2 hybrid
couplers) and phase shifters. To remove intersections of such components that degrade
the ability to extend the operation bandwidth, we designed a waveguide-based multilayer Butler matrix having two types of 3-dB coupler, one for the E-plane and one for the
H-plane. Therefore, our designed Butler matrix can be implemented without any
intersections. Each E-plane coupler connected by a dashed line covers two layers and isoperated by fitting the two layers together. The transmission waveguides labeled Ay (y=1, 2…8) are connected to the corresponding antenna ports that form the shape of the38

UCA on the top layer. Fig. VI-2 (b) shows our implemented 8×8 Butler matrix forming
the UCA that can operate over a 32-GHz bandwidth in sub-THz frequency bands.

(a)

(b)

Fig. VI-2. (a) Schematic design of 8×8 Butler matrix, (b) Implemented 8×8 Butler
matrix forming the UCA.
VI-3.2. Design of the Parabolic Reflector Antenna for Extending the Transmission
Distance
To extend the transmission distance of the OAM waves, we developed a parabolicreflector antenna. The antenna was designed for the plane wave source of OAM waves
using two paraboloidal surfaces with the same focal point and different focal lengths, as
shown in Fig. VI-3 (a) [3].

(a)

(b)

Fig. VI-3. (a) Parabolic reflector design for OAM multiplexing, (b) Photograph offabricated parabolic reflector antenna mounted with the UCA.
Because the focal point (F) is the same, a beam parallel to the axis of propagation (AS)from the feed antenna is bounced back by the two reflectors and becomes parallel again

39

(MM'), thereby expanding the beam diameter D’. This allows the aperture plane to beenlarged by the two reflectors. The focal length of the main reflector is 𝑓𝑚 , and that of𝑓

the sub-reflector is 𝑓𝑠 . The aperture D’ was determined to be 𝑑 × 𝑚 . That is, the aperture𝑓𝑠of the UCA can be enlarged by a factor of 𝑓𝑚 /𝑓𝑠 . Fig. VI-3 (b) shows our implemented
parabolic reflector antenna designed for 20-m transmissions using a UCA with 6-cm
diameter. The diameter of the main reflector was set to 35 cm to enlarge the aperture of
the UCA by a final factor of 5 using two reflectors. A fabrication margin of 5 cm wasprovided to account for a slight spread.
VI-4. Experimental Evaluations
We performed experiments transmitting OAM multiplexing signals using our designed
Butler matrix and parabolic reflector in sub-THz frequency bands, as shown in Fig. VI-4
(1). Table VI-1 summarizes our experimental parameters. The transmission signals ineach OAM mode were independently processed before and after the 8×8 Butler matrix
in the transmitter and receiver, respectively. Therefore, equalization between the signal
series in the OAM modes was not required. Fig. VI-4 (b) shows the representative
constellations of the transmitted signals in the 152–168 GHz band. We found that eight
different signals could be successfully transmitted using various modulation levels andchannel coding rates. Bit error rates of all the transmitted signals were less than 10-7.
Table VI-1. Experimental Parameters.
Parameters

ValuesParameters

ValuesOAM modes−4, −3, −2, −1, 0, 1, 2, 3

Total baud rate

16+16=32 GBd

Diameter of reflector

12 cm (Sub) / 35 cm (Main)

Frame length

32,400 symbols

Diameter of UCA

6 cm

Forward error
correction

BCH (Outer) / LDPC (Inner)

Num. of antennas

8 elements

Equalization

SC-FDE

Transmission distance

20 m

Modulation

16 QAM, 64 QAM

Frequency

136–152, 152–168 GHz

Coding rate

1/2, 3/5, 2/3, 3/4, 4/5, 5/6

Table VI-2 shows the experimental results. As a reference a 1-m distance experiment
was conducted using only UCAs without any parabolic reflectors. In both 1-m and-20 m
distance experiments, similar transmission rates larger than 700 Gb/s over the 32 GHz
bandwidth were achieved. Potential exists to further double the transmission capacity
by using polarization multiplexing to achieve over 1 Tb/s. We experimentally found a
total 1.4 Tb/s capacity using dual polarization at 1 m without a parabolic reflector
antenna and at 20 m with the parabolic reflector antennas, respectively.

40

(a)

(b)

Fig. VI-4. (a) Experimental environment for wireless OAM multiplexing transmissions
at 20 m in a shielded room, (b) Representative constellations of the transmitted signals
in the 152–168 GHz band.
Table VI-2. Experimental Results.
Frequency Band

Transmission rates (1 m)

Transmission rates (20 m)

Lower (136–152 GHz)

331 Gb/s

335 Gb/s

Higher (152–168 GHz)390 Gb/s

371.73 Gb/s

Total (136–168 GHz)

721 Gb/s

707.73 Gb/s

VI-5. Conclusion
To provide high-capacity wireless transmissions, we have been conducting research on
OAM multiplexing. Thus, we designed and implemented an 8×8 Butler matrix that
operates over a 32-GHz bandwidth to fully utilize a wide bandwidth in sub-THz
frequency bands. To extend the transmission distance, we also designed a parabolicreflector antenna for OAM multiplexing. We experimentally showed that wirelesstransmissions over 1.4 Tb/s are feasible using our Butler matrix and parabolic reflector
antenna.
REFERENCE[1] Recommendation ITU-R M.2160-0, Framework and overall objectives of the future
development

of

IMT

for

2030

and

beyond.

https://www.itu.int/dms_pubrec/itu-r/rec/m/R-REC-M.2160-0-202311-I!!PDF-E.pdf[2] H. Sasaki, Y. Yagi, R. Kudo, and D. Lee, “Demonstration of 1.44 Tbit/s OAM
multiplexing transmission in sub-THz bands,” in Proc. 2023 IEEE ICC Workshops,
2023.
[3] Y. Yagi, H. Sasaki, R. Kudo, and D. Lee, “Parabolic Reflector for UCA-based OAM
Multiplexing in sub-THz Band and Transmission Experiment,” in Proc. 2023 IEEE
GLOBECOM Workshops, 2023.

41

[4] N. Kamiya and E. Sasaki, “A Simple Closed-Loop Method for Compensating the
Impact of Antenna Misalignment in UCA-based OAM-MIMO Systems,” in Proc. 2023
IEEE ICC Workshops, 2023.

42

VII. Performance Analysis of OAM Multiplexing Using Parabolic Antenna
Shuhei SAITO†, Yasunori YAGI‡, Doohwan LEE‡, and Fumiaki MAEHARA†
Graduate School of Fundamental Science and Engineering, Waseda University†‡

NTT Network Innovation Laboratories, NTT Corporation

Abstract— Utilizing a parabolic antenna enhances the transmission performance oforbital angular momentum (OAM) multiplexing. This study investigates OAM
multiplexing transmission with a parabolic antenna for a link distance. We analyze thereceived power concerning link distance, carrier frequency, and receiving antenna radius,
as well as highlight the advantages of planar reception achieved by the parabolic
antenna. Additionally, we also demonstrate how OAM modes affect propagation
attenuation at different link distances.
VII-1. Introduction
The sixth-generation mobile communication systems (6G), which are projected tomaterialize by 2030, are expected to have a peak data rate of 1 Tb/s [1]. Improving smallcell backhaul and access link technologies is crucial [2]. Specifically, for small-cellbackhauls, line-of-sight wireless communication using fixed transmitter and receiverlocations is preferred over fiber-based solutions owing to its flexibility, scalability, andcost effectiveness [2]. Orbital angular momentum (OAM) multiplexing has garneredattention for its ability to simultaneously transmit multiple data streams through asingle aperture pair [3].
To enhance the multiplexing transmission performance of OAM, a method involving aparabolic antenna has been proposed and validated [4]–[6]. In [4] and [5], thetransmission performance of a particular OAM mode with parabolic antennas wasassessed through simulations and experiments. However, these studies did notinvestigate the effects of planar reception using parabolic antennas. Conversely, in [6],an analysis comparing parabolic and uniform circular array (UCA) antennas
demonstrated that propagation attenuation did not depend solely on the OAM mode.
However, this analysis focused on relatively short link distances. Thus, a comprehensive
evaluation of parabolic antennas across a wider range of link distances is warranted.
Considering the background presented above, we investigate the multiplexing
performance of OAM using parabolic antennas for a link distance [7]. In ourinvestigation, parabolic antennas are employed in both the transmitter and receiver toenhance transmission performance. We analyze the effectiveness of planar reception
facilitated by a parabolic antenna and compare it with that of UCA reception. This
analysis elucidates the effects of different OAM modes on propagation attenuation at
different link distances.

43

VII-2. Performance Analysis of OAM Multiplexing Using Parabolic Antenna
Fig. VII-1(a) shows the phase and intensity distributions of OAM signals for modes l= 1, 2, and 3. From this figure, a spiral phase with a uniform intensity distribution canbe confirmed regardless of l. Fig.

VII-1(b) illustrates OAM transmission using aparabolic antenna. The OAM signal was transmitted by reflecting it from the feed to thereceiver through the parabolic antenna. At the receiver, co-phase combination by aplanar antenna enhanced the received power, and the desired OAM mode can beextracted using the discrete Fourier transform. For shorter link distances, where beam
divergence was less pronounced, planar reception mitigated propagation attenuation,
particularly for higher OAM modes [6]. However, for longer link distances, severe beam
divergence reduced the effectiveness of planar reception achieved by the parabolic
antenna.
Fig. VII-2 shows the coordinates of the OAM signal transmission using a parabolic
antenna, which included the focal length fp, diameter Dp, depth hp, semi-open angle θp of
the parabolic antenna, and radius Rr for the receiving UCA or planar antenna.
Additionally, a circular microstrip antenna was used as the feed for the parabolic
antenna. In our analysis, we employed spherical coordinates (r′, θ′, φ') and rectangularcoordinates (x′, y′, z') at the transmitter, as well as (r, θ, φ) and (x, y, z) at the receiver.
The electric field of the reflected OAM signal on the transmit parabolic antenna canbe expressed as [6]

(

(

)

)

ˆ t' sin ' + ˆ'Et' cos 'Et' + ˆ'Et' exp ( − jk0 r' ) r' ,Er = Ei − 2 nˆ  Ei nˆ = − r'E(1)

where Ei is the electric field of OAM signal radiated from the feed and incident on theparabolic antenna; Et' and Et ' are the θ′ and φ' components of Ei , respectively; n̂ is thenormal unit vector of the parabolic antenna; r̂' , ˆ' , and ˆ' are the unit vectors of
l=3

0

x [m]

0.3

0.6-0.6

-0.3

0

x [m]

0.3

0.6-0.6

-0.3

0

x [m]

0.3

0.6

0.6

0

0.3

-10

Feed

DFT

-180

Planar antenna

Data

-120
-0.3

Data

-60

y [m]

OAM signal
-20

0

-30

-0.3
-0.6
-0.6

-0.3

0

x [m]

0.3

0.6-0.6

-0.3

0

x [m]

0.3

0.6-0.6

-0.3

0

x [m]

0.3

0.6

Short distanceIntensity [dB]

Intensity Distribution

0

-0.3

Parabolicreflector

IDFT

y [m]

60
0

Phase [deg]Phase Distribution

120

-0.6
-0.6

Lower OAM mode
⇒ Lower beam divergence

180

0.3

DFT

l=2

Data

l=1
0.6

Long distance
Low

Large

-40

Beam divergence

Effect of planar receiving

Higher OAM mode
⇒ Higher beam divergence
High

Small

(a) Signal distributions of OAM signals(b) OAM transmission with parabolic antenna
Fig. VII-1. Concept of OAM transmission.

44

hp

Dp

φ'

hp

#N

x

r'

θ'

z'

#q

θp

r

x'
O' y'

O

x

Feed

y

Dp

φ

θ

Rr

φ'

r'

z'

z

r

x'
O' y'

O

D

q

θ'

θp

y

Feed

φ

θ

R

Rr

z

D

fp

fp
Transmitter

Transmitter

Receiver

Receiver(a) UCA reception(b) Planar antenna reception
Fig. VII-2. Coordinates of OAM signal transmission employing parabolic antenna.
the spherical coordinates (r′, θ′, φ'); and k0 = 2π / λ denotes the wavenumber with thewavelength λ.
At the receiver, the electric field at point q of the receiving antenna can be expressedas [6]
Eq = ˆEq , + ˆ Eq , ,Eq , =
Eq , =

− jk0 f p e− j k0 ( r + f p + h p )2 r− jk0 f p e− j k0 ( r + f p + h p )

2 r

(2)
 ' 

j k0 f p sin  tan   cos(' + )
 ' 
2
(1 + cos  )  tan   ( Ex't cos  + E y't sin  ) e
dS ,
S
2
 

(3)

 ' 

j k0 f p sin  tan   cos(' + )
 ' 
2
(1 + cos  )  tan   ( − Ex't cos  + E y't sin  ) e
dS ,
S
2

Ex't = Et' cos ' − Et' sin ' , E y't = Et' sin ' + Et' cos ' ,

(4)
(5)

where S denotes the aperture of the parabolic antenna. Based on Eqs. (2)–(5), the received
power of OAM mode l with planar antenna reception, with consideration of co-phasecombination, can be calculated as
2

PPlanar,l = 

Rr

0

Rr
1 2
− jl
Eq e q d q dR = 

0
0
2

2

2

2

1 2
1 2
− jl
− jl
Eq , e q dq +Eq , e q d q dR . (6)


0
0
2
2

Meanwhile, the received power of OAM mode l with UCA reception can be obtained asshown in [7].
VII-3. Numerical Results
In this section, we assess the effectiveness of planar reception by comparing it with
UCA reception based on various parameters, including the link distance, carrier
frequency, and antenna radius. We evaluated the received power of the OAM modes (l =0, 1, ..., 4) at carrier frequencies of 80 and 300 GHz. We used a parabolic antenna with
an aperture diameter Dp of 0.6 m and a focal length fp of 0.4 m. For UCA reception, we
employed a receiving UCA with N = 16 antenna elements and assumed a negligible
element size.
Fig. VII-3 illustrates the received power versus receiving antenna radius R for two
different link distances (D = 40 and 400 m) at a carrier frequency of fc = 80 GHz. As

45

-100

fc = 80 GHz, D = 40 m

-150

l=0
l=1
l=2

l=3
l=4

-150

-200

-200
l=0
l=1
l=2

-250
0

fc = 80 GHz, D = 400 m

Received Power [dB]

Received Power [dB]

-100

l=3
l=4

0.2
0.4
0.6
0.8
Receiving Antenna Radius R [m]

-250
0

1

0.2
0.4
0.6
0.8
Receiving Antenna Radius R [m]1(a) D = 40 m(b) D = 400 m
Fig. VII-3. Received power vs. receiving antenna radius, where the carrier frequency
fc = 80 GHz.
-50

-50

fc = 300 GHz, Rr = 0.15 m
Planar Antenna
UCA
l=0
l=1
l=2

-100

Received Power [dB]

Received Power [dB]

fc = 80 GHz, Rr = 0.15 m
Planar Antenna
UCA

-100

l=0
l=1

-150

-150
l=2
l=3

l=3

l=4

-200
10

l=4

2

10
Link Distance D [m]

-200
10

3

10

102
Link Distance D [m]103(a) fc = 80 GHz(b) fc = 300 GHz
Fig. VII-4. Received power vs. link distance, where the radius of planar antenna or
UCA Rr = 0.15 m.
shown in Fig. VII-3(a), for OAM modes l = 2, 3, and 4, the received power decreased,
particularly for smaller R values. This reduction is due to the increasing severity of beam
divergence with higher OAM modes. Conversely, as shown in Fig. VII-3(b), as the link
distance D increased to 400 m, the effect of beam divergence was more significant thanthat of the case involving D = 40 m presented in Fig. VII-3(a), which resulted in a lower
received power for higher OAM modes regardless of R. Therefore, using a parabolic
antenna can mitigate the propagation attenuation, particularly for short link distances.
Fig. VII-4 depicts the received power versus the link distance D for each OAM mode lunder a fixed antenna radius (Rr = 0.15 m) for both the planar and UCA antennas. As
presented in Fig. VII-4, the planar antenna consistently yielded a higher received power
than the UCA, irrespective of the fc and D, owing to planar reception. Furthermore,
adopting planar reception uniformly reduced the propagation attenuation over short link
distances owing to the reduced beam divergence, particularly at high carrier frequencies
such as fc = 300 GHz. Conversely, at lower carrier frequencies or longer link distances,the propagation attenuation became mode-dependent regardless of the antenna type,owing to severe beam divergence.

46

VII-4. Conclusion
In this study, we examined the effect of link distance on OAM multiplexing
performance using a parabolic antenna. Our key finding was that planar reception
outperformed UCA reception for a link distance. The numerical results indicated that
adopting a planar antenna consistently improved the received power for all OAM modes
regardless of the link distance or carrier frequency, owing to its planar reception
properties. Shorter link distances exhibited reduced propagation attenuation owing to
the lower beam divergence, where OAM signals appeared primarily near the propagation
axis. Conversely, longer link distances experienced greater propagation attenuation evenwhen parabolic antennas were used, particularly at higher OAM modes, owing to severe
beam divergence.
REFERENCE[1] Z. Zhang et al., “6G wireless networks: Vision, requirements, architecture, and key
technologies,” IEEE Vehicular Technology Magazine, vol. 14, no. 3, pp. 28–41, Sept.
2019.
[2] Z. Pi, J. Choi, and R. Heath, “Millimeter-wave gigabit broadband evolution toward
5G: Fixed access and backhaul,” IEEE Communications Magazine, vol. 54, no. 4, pp.
138–144, Apr. 2016.
[3] D. Lee, H. Sasaki, H. Fukumoto, K. Hiraga, and T. Nakagawa, “Orbital angularmomentum multiplexing: An enabler of a new era of wireless communications,”
IEICE Transactions on Communications, vol. E100-B, no. 7, pp. 1044–1063, July
2017.
[4] T. Nguyen, R. Zenkyu, M. Hirabe, T. Maru, and E. Sasaki, “A study of orbital angularmomentum generated by parabolic reflector with circular array feed,” Proc. 2016
International Symposium on Antennas and Propagation (ISAP 2016), pp. 708–709,
Oct. 2016.
[5] Y. Yao, X. Liang, W. Zhu, J. Geng, and R. Jin, “Experiments of orbital angularmomentum phase properties for long-distance transmission,” IEEE Access, vol. 7, pp.
62689–62694, May 2019.
[6] Q. Wu, X. Jiang, and C. Zhang, “Attenuation of orbital angular momentum beamtransmission with a parabolic antenna,” IEEE Antennas and Wireless Propagation
Letters, vol. 20, no. 10, pp. 1849–1853, Oct. 2021.
[7] S. Saito, Y. Yagi, D. Lee, and F. Maehara, “Link distance and carrier frequency
dependence of propagation attenuation in OAM multiplexing using parabolic
antenna,” Proc. 2023 IEEE Annual International Symposium on Personal, Indoor andMobile Radio Communications (PIMRC 2023), pp. 1–5, Sept. 2023.

47

VIII. Antenna Array Module for Sub-THz Communications
Shiro Ozaki, Fujitsu Limited
Takuichi Hirano, Tokyo City University

Abstract—For sub-THz communication applications, we developed an antenna array
module with element pitch less than 1 wavelength (1λ) at 300 GHz. The 1 × 4 waveguidebased antenna array module showed an antenna gain of 10 dBi and a beam steering
angle of ±10° with reduced grating lobes.
VIII-1. Introduction
Sub-THz bands, particularly the 300-GHz band, are considered as one of candidates
for next-generation wireless communications [1]. For the sub-THz indoor use case shownin Fig. VIII-1 [2], it is essential for the access point to perform beamforming withantenna arrays. The large grating lobes have been a critical issue to realize practicalantenna arrays. This is because the size of active devices such as power amplifiers (PAs)tend to be large for the sub-THz wavelengths (λs), i.e. 1 mm at 300 GHz [3]. In this study,
we demonstrated an antenna array module with element pitch less than 1λ for reducing
grating lobes.

Fig. VIII-1. Indoor use case of 300-GHz band wireless communications (Reprinted
with permission from [2], copyright 2023 IEEE)

48

VIII-2. Demonstration of 1 × 4 Antenna Array
Fig.

VIII-2 [2] illustrates the proposed 1 × 4 antenna array module. 4-element

waveguide antennas and PAs are integrated heterogeneously in the module. Sub-THz
signals propagate on microstrip lines (MSLs) formed on a printed substrate. The PAs aremounted on the printed substrate by flip chip assembly. In addition to these technologies
for heterogeneous integration, we are developing PA monolithic microwave integrated
circuit using InP-based metal-oxide-semiconductor high-electron-mobility transistors(MOS-HEMTs) [4, 5] to achieve high equivalent isotropic radiation power.
Herein, a waveguide-based antenna array (without PAs) was fabricated with anelement pitch of only 0.7 mm. In Fig. VIII-3(a) [2], the aperture area of each waveguide
antenna was 0.6 mm × 0.432 mm. The thickness of metal walls between the aperturearea was designed to be 0.1 mm to achieve the element pitch of 0.7 mm. Polyimide wasused as a printed substrate by considering low transmission loss (low dielectric constantand low loss tangent) and a design rule for heterogeneous integration with PAs: highdensity through-hole vias for ground pads used in flip chip mounting. To suppress
radiation from the substrate at 300 GHz, the pitch of ground vias are required less than
0.1 mm (0.1λ) [6]. At the MSL-Waveguide transition, the width of the polyimide
substrate was designed to be 0.2 mm in the floating state (Fig. VIII-3(b) [2]). In this case,the thickness of the polyimide substrate and the width of the signal line formed on thesubstrate were 25 and 50 µm, respectively to be the characteristic impedance of 50 Ω.

Fig. VIII-2. Proposed 1 × 4 antenna array module with PAs (Reprinted withpermission from [2], copyright 2023 IEEE)49

Fig. VIII-3. Photograph of the developed 1 × 4 antenna array module (Reprinted
with permission from [2], copyright 2023 IEEE)
The performance of the developed 1 × 4 antenna array module is shown in Fig. VIII-4[2]. The developed antenna array achieved a gain of 10 dBi and a beam steering angle of
10° by varying the phase difference between each port from 0° to 45° (Φ1 = 0°, Φ2 = 45°,
Φ3 = 90°, Φ4 = 135°), which is in good agreement with electro-magnetic simulation.
Moreover, simulations revealed that the beam can be steered by ±24° by varying thephase difference between each port from 0° to 120° (Φ1 = 0°, Φ2 = ±120°, Φ3 = ±240°,
Φ4 = ±360°), as shown in Fig. VIII-5 [2]. This phase shift reduced the antenna gain from
10.6 to 9.35 dBi and increased the grating lobe level to 1.95 dB at ±54°. By reducing theelement pitch to be less than 1 mm, the grating lobes of the antenna array module weresuccessfully reduced compared to previous report [7] that used horn antennas with anelement pitch of 1.25 mm.

50

Fig. VIII-4. Performance of the developed 1 × 4 antenna array module (Reprinted withpermission from [2], copyright 2023 IEEE)

Fig. VIII-5. Performance of the developed 1 × 4 antenna array module (Reprinted withpermission from [2], copyright 2023 IEEE)

51

VIII-3. Conclusion
Herein, an antenna array module with element pitch less than 1λ at 300 GHz wasdeveloped for sub-THz communication applications. The 1 × 4 waveguide-based antenna
array module exhibited a gain of 10 dBi and a beam steering angle of ±10°. By electromagnetic simulation, it was confirmed that the beam can be steered by ±24° by varying
the phase difference between each port from 0° to ±120°.
VIII-4. Acknowledgements
This work was partially supported by "R&D on THz band ultra high-capacity wirelesscommunications for beyond 5G (No.00301)" from the commissioned research by National
Institute of Information and Communications Technology (NICT), and "The research anddevelopment project for the expansion of radio spectrum resources (JPJ000254)" of the
Ministry of Internal Affairs and Communication, Japan.
REFERENCE[1] V. Petrov et al., “First Standardization Efforts for Sub-Terahertz Band
Communications toward 6G,” IEEE Communications Magazine, vol. 58, no. 11, pp.
28-33, Nov. 2020.
[2] S. Ozaki et al., “A Waveguide Antenna Array Module with Controlled Element Pitch< 1λ for Sub-THz Communications,” CAMA, pp.281-284, Nov. 2023.
[3] T. Hirano, “Series-Fed Printed-Dipoles With a Director for Substrate Lateral
Radiation,” ISAP, pp.537-538, Oct. 2022.
[4] Y. Kumazaki et al., “High-Efficiency 250-320 GHz Power Amplifiers Using InP-Based
Metal-Oxide-Semiconductor

High-Electron-Mobility

Transistors,”

IEICE

Transactions on Electronics, 2023MMP0005, Aug. 2023.
[5] S. Ozaki et al., “Improved fT/fmax in wide bias range by steam-annealed ultrathinAl2O3 gate dielectrics for InP-based high-electron-mobility transistors,” Appl. Phys.
Express 15, 041001, Mar. 2022.
[6] Y. Kawano et al., “Flip chip assembly for sub-millimeter wave amplifier MMIC on
polyimide substrate,” MTT-S, pp.1-4, Jun. 2014.
[7] S. Rey et al., “A Phased Array Antenna with Horn Elements for 300 GHz
Communications,” ISAP, pp.122-123, Oct. 2016.

52

IX. Virtualized terminal technologies for high-capacity communications in
Beyond 5G era
Satoshi Ito, KDDI Research, Inc.
Yoshio Kunisawa, KDDI Research, Inc.
Takahiro Hayashi, KDDI Research, Inc.
Yoshiki Sugimoto, Nagoya Institute of Technology
Kunio Sakakibara, Nagoya Institute of Technology
Nobuyoshi Kikuma, Nagoya Institute of Technology

Abstract— In anticipation of the Cyber Physical Systems (CPSs) integration in 2023,
this paper presents a "virtualized terminal" concept for enhanced uplink capacity inbeyond 5G networks to overcome the space and power constraints of conventional MIMO
systems. Operating in the subterahertz band, the proposed architecture includes
offloading antenna functions to relay devices (RDs), thus enabling high-capacity datatransmission. The experimental results show a 19.2 Gbps transmission rate. Future
work will seek to achieve the Beyond 5G target of 100 Gbps through advanced component
miniaturization and leveraging a network of five RDs.
IX-1. Introduction
As 2030 approaches, the integration of physical and cyber spaces, which is known as
Cyber Physical System (CPS), is expected [1]. Upon realizing the broad integration a
CPS, there will be an increased demand for uploading large amounts of data to networks.
Consequently, there will be a greater need for high-capacity uplink transmission
capabilities than ever before. MIMO is considered to be a method for enhancing uplink
capacity, but challenges such as the space required for antenna installation and thepower consumption of amplifiers limit the increase in the number of antennas that canbe achieved within devices. Therefore, the present study explores the concept of a“virtualized terminal” that relocates the antenna functionality within a device tosurrounding wireless devices to communicate with base stations, thereby enabling highcapacity uplink communication [2]. This paper presents the conceptual and architectural
overview such a virtualized terminal, including utilization of the subterahertz band, andreports the current status of proof-of-concept experiments for virtualized terminals.
IX-2. Virtualized terminal technologies
The proposed virtualized terminal is shown in Fig. IX-1. This virtualized terminal
consists of user equipment (UE) with which the user directly interacts with, as well as
multiple relay devices (RDs), which are installed in wearable devices. This configuration
enables spatial multiplexing transmission with many antennas without constraint from53

the installation space or power consumption within the UE. The communication between
the virtualized terminal and the base stations is assumed to utilize the 40 GHz band
(37.0 GHz to 43.5 GHz), which is being considered for allocation to 5G in the Frequency
Reorganization Action Plan of Japan [3].
The virtualized terminal is intended to achieve a data rate of 100 Gbps, which isanticipated for use in Beyond 5G [1] applications, by transmitting 10 streams using a
QPSK modulated signal with a 4.8 GHz bandwidth to five RDs that support orthogonal
dual polarization multiplexing. However, due to interference concerns, it is difficult to
use the millimeter-wave band simultaneously for communication between the UE andthe RDs, which is also used for communication with the base stations. Therefore, thevirtualized terminal uses different frequency bands for wireless transmission from the
UE to each RD in the subterahertz band allocated to land mobile services by the ITU-R,
ranging from 252 GHz to 296 GHz [4]. While a total bandwidth of 24 GHz is required forthe five RDs, the abovementioned subterahertz band can potentially provide a 44 GHz
bandwidth, which is a sufficient capacity even when accounting for guard bands.
Moreover, large path loss is a critical issue in subterahertz band communication.
However, the expected propagation distance in the virtualized terminal configuration isapproximately 1 m in the subterahertz band, and it is anticipated that frequency
conversion to the millimeter-wave band will enable communications over distances ofmore than 100 m to the base station [5]. An example of the link budget design is shownin Fig. IX-2. The transmission power for the subterahertz band is based on values fromprevious research, while the millimeter-wave band uses values equivalent to 5G, with asimple model using free space path loss applied. In this example, transmission from the
UE to the RD over a distance of 0.9 m in the subterahertz band and subsequenttransmission over a distance of 100 m to the base station in the millimeter-wave bandusing QPSK are both feasible.
Fig. IX-3 shows an example of the configuration of the virtualized terminal. In the UE,
baseband signals for each RD are converted to different frequencies in the subterahertzband by the subterahertz band front-end component. Signals in each subterahertz band
are radiated toward each RD by a multibeam antenna. The RDs convert the received
subterahertz band signals from the UE into frequencies used in the millimeter-wave
band for communication with the base station and then radiate these signals from themillimeter-wave band antennas. By using all RDs to emit radio waves at the same
frequency in the millimeter-wave band toward the base station, the virtualized terminal
is configured with multiple antennas realized by the RDs.

54

Fig. IX-1. Concept of a virtualized terminal.

Fig. IX-2. Link budget for the virtualized terminal.

Fig. IX-3. Structure of the virtualized terminal.
IX-3. Demonstration experiment
To evaluate a wireless communication system using a virtualized terminal, a proof-ofconcept experiment was conducted with a measurement system comprising two
transmitters and two receivers in the subterahertz band and a multibeam lens antenna
[6].

55

Fig. IX-4 shows the measurement system used for the proof-of-concept experiment.
An arbitrary waveform generator (AWG) was used to generate a 4.8 GHz bandwidth
OFDM signal with a center frequency of 37.5 GHz, modulated with two different data
sequences, which was subsequently input into two subterahertz band transmitters. The
distance between the subterahertz band transmitting and receiving antennas was set to
20 cm for channel 1 and 33 cm for channel 2. The modulated signals from thetransmitters were received by two receivers via a subterahertz band multibeam antenna,frequency-converted to the millimeter-wave band, level-adjusted, and subsequently fed
to the vertical and horizontal polarization ports of an orthogonal dual-polarization
antenna for radiation in the millimeter-wave band. On the base station side, the signals
received by the dual-polarization antenna were amplified by a low-noise amplifier (LNA)and subsequently input into two channels of an oscilloscope for vertical and horizontal
polarization. After demodulating the OFDM signal, the bit error rate (BER) was
calculated from the demodulated QPSK signals. The results were subsequently used to
calculate and evaluate the physical transmission rate for the 4.8 GHz bandwidth.
Fig. IX-4 also shows the constellation and power spectrum of the received signal whenthe millimeter-wave band propagation distance is 10 m. As shown in the figure, thephysical transmission rate for both channel 1 and channel 2 was approximately 9.6 Gbps,
for a combined total of 19.2 Gbps. This finding indicated that with five RDs supporting
dual polarization, it is feasible to realize the 100 Gbps equivalent required for Beyond
5G. However, the received signal exhibited a difference of more than 10 dB within thesignal transmission band, and the constellation was dispersed. In this demonstration,
discrete RF components were used, but it is expected that the use of MMIC (Monolithic
Microwave IC) will lead to an improvement to the in-band frequency characteristics.

Fig. IX-4. Experimental setup.

56

IX-4. Conclusion
This paper reports the conceptual and architectural overview of a virtualized terminal
utilizing the subterahertz band to achieve high-capacity uplink communication. The
data confirmed that by frequency-converting subterahertz band QPSK signals to themillimeter-wave band using RDs and wirelessly relaying these signals through twosystems, a physical transmission rate of 19.2 Gbps was achieved, as calculated from thebit error rate (BER) at a millimeter-wave transmission distance of 10 m. Futureadvancements to this work will continue the miniaturization and enhancement ofsubterahertz band RF components by introducing MMIC technology. This improvement
is to facilitate a physical transmission rate of 100 Gbps by implementing wirelesstransmission through ten channels using a configuration of five RDs. Therefore, theproposed system is intended to achieve a physical transmission rate equivalent to the
100 Gbps by conducting wireless transmission across ten systems using five RDs.
IX-5. Acknowledgements
These

research

results

wereobtained

from

the

commissioned

research(JPJ012368C00401) by National Institute of Information and Communications
Technology (NICT), Japan.
REFERENCE[1] Beyond 5G Promotion Consortium White Paper Subcommittee, “Beyond 5G White
Paper,” Mar. 2023, https://b5g.jp/en/output/[2] KDDI, KDDI Research, Inc., “Beyond 5G/6G White Paper ver.2.0.1,” Oct. 2021,
https://www.kddi-research.jp/english/tech/whitepaper_b5g_6g/[3] Ministry of Internal Affairs and Communications Japan，“Solicitation of Opinions on
Draft

Frequency

Reorganization

ActionPlan(Fiscal2022

Version),”https://www.soumu.go.jp/main_sosiki/joho_tsusin/eng/pressrelease/2022/9/01_03.htm
l[4]

ITU-R,“WorldRadiocommunication

Conference

2019

Final

Acts,”

https://www.itu.int/dms_pub/itu-r/opb/act/R-ACT-WRC.14-2019-PDF-E.pdf[5] Y. Kunisawa and Y. Amano, “Virtualized terminal utilizing terahertz band radio
waves for Beyond 5G : Link budget analysis,” IEEE VTC2022-Spring, June. 19-22,
2022. DOI: 10.1109/VTC2022-Spring54318.2022.9860515[6] T. Sugiyama, Y. Sugimoto, K. Sakakibara and N. Kikuma, “Low-profile Multi-beamLens Antenna with Corrugation to Suppress Surface Reflection of High-permittivity
Material in 300 GHz Band,” 2023 IEEE International Symposium on Antennas andPropagation, Oct. 2023, Kuala Lumpur, Malaysia.

57

X. Sub-THz multi-beam antennas for virtualized terminal technologies
Kunio Sakakibara, Nagoya Institute of Technology
Yoshiki Sugimoto, Nagoya Institute of Technology
Satoshi Ito, KDDI Research, Inc.
Yoshio Kunisawa, KDDI Research, Inc.
Takahiro Hayashi, KDDI Research, Inc.
Nobuyoshi Kikuma, Nagoya Institute of Technology

Abstract—Various multi-beam antennas are in the development at Sub-THz band.
Plano-convex lenses are potentially capable to design in lower profile. Spherical lenses
essentially never generate scan loss. Rotman lens arrays are low-profile planar antennas.
These multi-beam antennas are proposed to apply to the virtualized terminal
technologies of user-centric architecture for future beyond 5G/6G systems.
X-1. Introduction
The sub-terahertz band is attracting attention for next-generation mobile
communication beyond 5G/6G. Developments are strongly driven in wide range oftechnical fields including antennas and propagation. However, due to its highpropagation loss of sub-terahertz waves, it is not easy to apply MIMO technologies which
utilizes in multipath environments. One possible way to utilize sub-terahertz
communication can be a virtualized terminal technology [1]. This report demonstrates
the antenna performances required for the virtualized terminal technologies andpresents the status of multi-beam antennas currently under development.
X-2. Required function of Sub-THz antennas for virtualized terminal technologies
A user-centric architecture has been proposed to improve the communication quality
and to extend the service area for future cellular systems [2]. MIMO communication
technologies are widely used in various wireless communication systems such as cellular
systems and WiFi. MIMO antennas in a user terminal are extended out to thesurrounding area around the user terminal in the virtualized terminals of a user-centricarchitecture as shown in Fig. X-1. The user terminal and the relay devices are connectedby high-speed sub-terahertz communications with narrow-beam antennas.
There are multiple relay devices for one user terminal. Therefore, as shown in Fig. X-2,
a multi-beam antenna with independent beam-scanning is required for the user terminal.
On the other hand, relay devices require a beam scanning function in a single beam.
Various beam scanning antennas have been proposed in general. Low-profile planarantennas are advantageous for installation in user terminals and relay devices. However,the feeding loss of the transmission line is serious in the sub-terahertz band. Lens

58

antennas perform very well. However, the focal length and the lens thickness cannot be
avoided essentially. Lens antennas can still be candidates because high gain is obtainedeven with a small size.
Mechanical scanning using a lens antenna and phased array using a planar antenna
are possible. However, we decided to develop a dielectric lens multi-beam antenna foruser terminals and a planar multi-beam antenna with a Rotman lens for relay devices.

Fig. X-1. Virtualized terminal technologies
for user-centric architectureFig.

X-2. Simultaneous communication
between a user terminal and relay devices

X-3. Plano-convex dielectric lens multi-beam antennas
A lens antenna can scan the beam by shifting the position of the primary radiator fromthe focal point of the lens. A multi-beam antenna was developed by arranging eight hornantennas in an arc around the focal point of a plano-convex polypropylene lens (r = 2.4)
with a diameter of 10mm as shown in Fig. X-3(a). The radiation patterns are presented
in Fig. X-3(b). Eight beams with a pitch of approximately 7.5 degrees cover ±30 degrees
with a gain higher than 22.8dBi [3].

Fig. X-3 (a) Developed plano-convex multi-beam lens antenna.

59

Fig. X-3 (b) Multi-beam radiation patterns and gains of the plano-convex lensantenna.
X-4. Spherical dielectric lens multi-beam antennas
Plano-convex lenses have the problem of scan loss due to the reduction of projected
aperture size and the spill over. Since the Luneberg lens has a point-symmetric structure,scan loss is not generated in principle [4]. However, it requires a layered structure in theradial direction, which is not practical in the sub-terahertz band due to manufacturing
constraints. Therefore, we developed a spherical lens multi-beam antenna. Although thefocusing performance is degraded, the manufacturing is practical. A multi-beam antenna
is formed by arranging 16 horn antennas around the focus of the polypropylene sphericallens with a diameter of 20mm as shown in Fig. X-4(a). The multi-beam radiation
patterns with the gains are presented in Fig. X-4(b). Peak gain higher than 30dBi was
obtained. Sixteen beams with a pitch of approximately 4 degrees cover ±30 degrees witha gain higher than 26.7dBi [5].

Fig. X-4. (a) Spherical lens multi-beam antenna.

60

Fig. X-4 (b) Multi-beam radiation patterns and gains of the spherical lens antenna.
X-5. Rotman lens multi-beam antennas
Planar antennas have the advantage of low-profile. A compact multi-beam antenna
was designed to compose Rotman-lens beam-forming circuit on the back of the microstrip
array antennas as shown in Fig. X-5(a). The antenna can be fabricated in multi-layer
substrate. Two transmission line transitions were designed to connect the components.
The transition between substrate-integrated waveguide (SIW) and microstrip line isused to connect the microstrip array and the Rotman lens. The transition between the
waveguide and SIW is used at the feeding from the back. As a result of electromagnetic
simulation, five-beam characteristics with a 12-degree pitch was confirmed as shown in
Fig. X-5(b) [6].

(b) Multi-beam radiation patterns(a) Configuration

Fig. X-5. Rotman-lens multi-beam planar antenna.
X-6. Conclusion
We are in the development of multi-beam antennas for virtualized terminal
technologies of user-centric architecture expected for future beyond 5G/6G. We continue

61our development efforts to lower the profile of lens antennas and to improve efficiency ofplanar antennas.
X-7. Acknowledgements
These

research

results

wereobtained

from

the

commissioned

research(JPJ012368C00401) by National Institute of Information and Communications
Technology (NICT), Japan.
REFERENCE[1] KDDI

Corporation, KDDI Research, Inc., “Beyond 5G/6G whitepaper,” Mar. 2021.

[2] K. Yamazaki, T. Ohseki, Y. Amano, H. Shinbo, T. Murakami and Y. Kishi, "Proposal
for a User-Centric Ran Architecture Towards Beyond 5G," 2021 ITU Kaleidoscope:
Connecting Physical and Virtual Worlds (ITU K), Geneva, Switzerland, 2021, pp. 17.
[3] T. Sugiyama, Y. Sugimoto, K. Sakakibara and N. Kikuma, “Low-profile Multi-beamLens Antenna with Corrugation to Suppress Surface Reflection of High-permittivity
Material in 300 GHz Band,” 2023 IEEE International Symposium on Antennas andPropagation, Oct. 2023, Kuala Lumpur, Malaysia.
[4] G. Peeler and H. Coleman, "Microwave stepped-index luneberg lenses," in IRE
Transactions on Antennas and Propagation, vol. 6, no. 2, pp. 202-207, April 1958, doi:
10.1109/TAP.1958.1144575.
[5] T. Tsuchida, Y. Sugimoto, K. Sakakibara and N. Kikuma, “High Crossover Level
Design of a 300 GHz Band Multi-beam Spherical Homogeneous Dielectric Lens
Antenna,” IEICE Tech. Rep., vol. 123, no. 223, AP2023-93, pp. 27-31, Oct. 2023.
[6] S. Kishi, K. Sakakibara, Y. Sugimoto, T. Morioka, J. Kim and N. Kikuma, “274-GHz
Substrate-Integrated-Waveguide Fed Rotman-Lens Multibeam Antenna Integrated
in Multi-Layer Substrate,” ISAP2023, RS28.4, Nov. 2, 2023.

62

XI. Sub-THz band device technologies
Yasunori Suzuki, NTT DOCOMO, INC.
Hiroyuki Takahashi, NTT Device Technology Labs., NTT Corporation
Toshihiro Ohki, Shiro Ozaki, Yusuke Kumazaki, Fujitsu Limited

Abstract—This chapter presents sub-THz band device technologies from high powertransistors to RF components. The research target of sub-THz band device technologies
is identified for sub-THz band mobile communication system in the future. The validityof the presented sub-THz band device technologies confirms the experiment andsimulation results.
XI-1. Introduction
Mobile communication systems have continuously evolved for offering various services.
The fifth generation (5G) mobile communication system has been launched from 2020,
while 5G mobile communication system supports millimeter-wave band operation forhigher data transmission.
For providing extremely higher data transmission in the future, many research groups
focuses on sub-THz band which has enough bandwidth for this purpose. The researchtarget of sub-THz band operation shows the throughout of more than 100 Gbit/s [1]. The
transmission distance from base station to terminal needs more than 100 m withcorresponding to current service areas. However, the current status is basic research
stage. There are many technical issues for exploring sub-THz band operation on mobile
communications.
First, RF circuit technologies should be revealed for achieving the research target of
sub-THz band operation. RF core technologies are presented for improving quality ofdata transmission. One is a frequency domain equalizer for providing widely flat
frequency response of transmitter. The other is a linearizer for compensating non-linear
components generated by power amplifier. This chapter presents the research results ofthese technologies in detail.
Next, power amplifier technologies are presented. There is the trade-off between
frequency and output power / efficiency from the operational principle of transistor.
Therefore, it is not sufficient the output power and efficiency of sub-THz band power
amplifier in the current status. According to proving various use-case scenarios by subTHz band mobile communications, it requires higher output power of sub-THz band
power amplifier for covering service area. Furthermore, the efficiency of sub-THz band
power amplifier such as more than 100 GHz becomes low level, while the power
consumption of one seems to become extremely larger level. This chapter shows the highoutput power and high efficiency techniques for 100 GHz and 300 GHz power amplifiers.

63

Finally, RF device technologies are presented for high quality of transmission at 300
GHz band. One is the mixer for operating wider bandwidth, which can realize for 100
Gbit/s transmission at 300 GHz band. The other is the front-end IC which can attain the
beamforming function at 300 GHz band. This chapter shows the experimental results.
XI-2. Frequency Domain Equalization Technology
It is well-known that the frequency response of transmitter for providing wider
bandwidth transmission is very important role for offering higher quality of transmission.
The bandwidth of sub-THz band transmitter will seem to be wider bandwidth than thatof cellular-band transmitter. However, the frequency response of actual sub-THz banddevice is difficult to become widely flat. We proposed the frequency domain equalizer forproviding widely flat frequency response [2].
Fig. XI-1(a) and (b) show the measurement configuration and experimental results ofthe proposed equalizer, respectively. The transmitter shown in Fig. XI-1(a) consists of
the arbitrary waveform generator (AWG), up-converter, filter, amplifier, coupler, local
oscillator, down-converter, and oscilloscope. The receiver shown in Fig. XI-1(b) is theanalyzer. The inverse of frequency response parameter is calculated in the oscillator
using the down-converted signal from the output signal of power amplifier, while theproposed equalizer using the parameter is set in the inside of AWG.

(a)(b)
Fig. XI-1. (a) Measurement configuration and (b) experimental results.
(Roll-off factor: 0.22, Modulation: 64QAM, IF: 8.5 GHz, RF: 155 GHz)
According to Fig. XI-1(b), the proposed equalizer can sufficiently improve the error
vector magnitude (EVM) results. The EVM improve about 13 dB at the baud rate of 8
Gbaud. In this case, the data transmission becomes about 48 Gbit/s. From these results,
the proposed equalizer is useful for providing sufficient EVM performance.

64

XI-3. Non-Linear Components Compensation Technology
In general, non-linear distortion components compensation is very important for
improving transmission quality such as bit error rate performance and EVM
performance. The millimeter-wave-band base station usually uses digital distortion
linearizer for compensating the non-linear distortion components. The sub-THz bandbase station will seem to employ linearization technology for the same purpose of themillimeter-wave-band base station. In addition, the linearization technology should
provide low power consumption for compact-size sub-THz band base stations. From thesereasons, we proposed an analogue predistorter operating sub-THz band [3]. Fig. XI-2(a)shows the basic configuration of sub-THz band transmitter, while the proposed
predistorter is set in front of power amplifier operating sub-THz band.
Fig. XI-2(b) shows the computer simulation results employing the InP HBT model [4]at 100 GHz. The test signals are two CWs with the same amplitude and difference
frequencies at 100.0 GHz and 100.1 GHz. The IM3 means the power of third-orderintermodulation distortion components. From Fig. XI-2(b), the proposed predistorter can
reduce IM3 component about 10 dB at the same input power per tone. In general,analogue predistorter does not consume power because of using a diode in IC. Therefore,
the sub-THz band transmitter does not pay attention to increase power consumption byusing the proposed predistorter. There are still technical issues for realizing the proposed
predistorter.

(a)(b)
Fig. XI-2. (a) Basic configuration and (b) simulation results.
XI-4. Device technology for 100 GHz band power amplifier
GaN-based high-electron-mobility transistors (HEMTs) have advantages in highpower operation because of their high critical electric fields. In addition, a high maximum
oscillation frequency enables the amplification of 100 GHz signals. In order to realize
further high frequency operation, insertion of AlN spacer layer between barrier andchannel layers is effective in improving electron mobility, but can degrade contact
resistance, in general.

65

In this section, we mentioned a low-resistance and low-thermal-budget ohmic contact
by introducing periodic microstructures for AlGaN/AlN/GaN HEMTs [5]. An ohmic
contact with periodic microstructures was developed through low-damage patternedrecess etching and metallization via low-temperature annealing (600˚C) as shown in Fig.
XI-3(a). The optimization of structural parameters of periodic microstructures allowed a
low contact resistance of 0.29 mm, which is less than half of that of the conventional
planar-recessed ohmic contact prepared in same study. A simplified equivalent circuit
model reproduced the experimental results and predicted that enhanced metal/channel
direct contact areas contributed to the reduction of contact resistance.
Pulsed current–voltage measurements revealed that an AlGaN/AlN/GaN HEMT withperiodic microstructures showed considerably suppressed current collapse comparedwith that of a selective-area growth (SAG) ohmic contact, indicating that the
introduction of trap states was suppressed in periodic-structured devices owing to lowthermal-budget processes. Under high off-stress conditions, periodic-structured devices
showed the highest drain current at knee voltage compared to the devices with aconventional planar-recessed ohmic contact and an SAG ohmic contact as shown in Fig.
XI-3(b). Therefore, ohmic contact with periodic microstructures is one of the best
techniques to achieve low contact resistance with low-thermal-budget processes.

Fig. XI-3. (a) Schematic cross-section of the GaN HEMT with periodic microstructuresand (b) correlation between drain current at knee voltage and quiescent drain voltage
obtained by pulsed Id-Vds measurement [5]XI-5. Device technology for 300 GHz band power amplifier
InP-based HEMTs with InGaAs channel and InAlAs electron supply layers show
higher output power characteristics in the 300 GHz band compared with other materials.
However, a large gate leakage current and a low breakdown voltage of them limit theiroperations. To address these issues, metal-oxide-semiconductor (MOS) gate structureswith a low gate leakage current are desirable for radiofrequency (RF) operation across awide bias range. Among the various types of dielectrics used to improve the performance66

of oxide-gate HEMTs, Al2O3 has been one of the most attractive because of its wide bandgap (∼7.0 eV), high dielectric constant (∼9.0), and high breakdown voltage (∼10 MV cm−1).
Furthermore, considering the tradeoff between effective reduction in gate leakage
current and degradation of transconductance (gm), it is necessary to optimize thethickness of oxide film. Thus, atomic layer deposited (ALD)- Al2O3 is suitable for highfrequency insulated-gate HEMTs.
Fig. XI-4(a) shows the schematic illustration of the InP-based MOS HEMTs withultrathin- Al2O3 gate dielectrics [6]. After the gate recess process, 2 nm thick Al2O3 films
were deposited on the recess structure using the remote plasma ALD method at 300 °C.
Trimethylaluminum (TMA) and O2 plasma were used as precursors, where O2 plasma
was used for increasing the density and decreasing the leakage current of ALD- Al2O3.
To minimize the defects in Al2O3 films, we applied steam annealing at 300°C for 1 h postdeposition. According to previous studies, annealing in an oxygen-containing
environment enhances the passivation of dangling bonds, including oxygen vacancies inoxide films. Moreover, it is expected that H2O vapor (steam) accelerates the hydrolysis
of carbon impurities in Al2O3 films caused by TMA.
Fig. XI-4(b) and (c) show the three-terminal drain and gate current versus gate voltage(Id, Ig-Vgs) and small signal characteristics of InP-based MOS and Schottky-gate HEMTs,
respectively. High fmax of > 600 GHz and extremely flat fmax were successfully achieved
across a wide bias range on InP-based HEMTs with a steam-annealed ultrathin- Al2O3
gate dielectric, due to a reduction in gate leakage current and an increase in forward
breakdown voltage.
Furthermore, a broadband high-efficiency six-stage common-source PA MMIC wasdeveloped using InP-based MOS HEMTs [7]. A small-signal gain greater than 15 dB was
achieved in 242-322 GHz, corresponding to 80 GHz bandwidth as shown in Fig. XI-5(a).
The input-output power (Pin-Pout) characteristics of the fabricated PA MMIC at 260 GHz
are illustrated in Fig. XI-5(b). Although the saturated output power (Psat) and peak PAE
could not be evaluated due to the output shortage of a signal source, the fabricated PA
MMIC demonstrated a high PAE greater than 7.0%, high drain efficiency greater than
7.2%, and high output power greater than 10.6 dBm, corresponding to an output power
density of greater than 290 mW/mm at the final stage. To our knowledge, the PA MMIC
developed in this work achieved the highest PAE among all the J-band solid-state PAs.

67

Fig. XI-4. (a) Schematic cross-section, (b) Id-Vgs characteristics and (c) fmax calculated
from S-parameters of the InP-based MOS-HEMT compared with Schottky gate HEMT(Reproduced from ref. 6, copyright 2017 by the Japan Society of Applied Physics).

Fig. XI-5. (a) S-parameters and (b) power characteristics of PA MMIC with InP-based
MOS HEMTs [7]XI-6. Challenge of utilizing over 100 Gbit/s per channel
Standardization of the 300-GHz band with a channel bandwidth of over 25 GHz has
been discussed in IEEE 802.15.3d [8]. The use of such a broad channel bandwidth ispromising for high-speed wireless systems with data capacities of over 100 Gbit/s perchannel. Given these requirements, an extremely wide operating frequency band isneeded for radio-frequency devices in the 300-GHz band. Subsequently, over 100-Gbit/s
transmission using multiple channels has been reported [9]. The 300-GHz-band
fundamental mixer IC shown in Fig. XI-6(a) has an operating band of 100 GHz ranging
from 220 to 320 GHz. In addition, back-to-back data transmission using the WR3.4
waveguide mixer module shown in Fig.

XI-6(b) was performed in three different

frequency bands. The maximum data rates of the 1st band (center frequency (fc) of 227GHz), 2nd band (fc of 257 GHz), and 3rd band (fc of 307 GHz) are 120 Gbit/s (20 GBaud
64QAM), 168 Gbit/s (42 GBaud 16QAM), and 152 Gbit/s (38 GBaud 16QAM),

68

respectively. In addition, dual-channel back-to-back data transmission shown in Fig.
XI-6(c) was also demonstrated using two mixer modules for a transmitter and a receiver.
The maximum data rate of the high band (fc of 316.5 GHz) and the low band (fc of 261
GHz) are 112 Gbit/s (28 GBaud 16QAM) and 128 Gbit/s (32 GBaud 16QAM), respectively.
The total data rate of a simultaneous signal transmission was 240 Gbit/s.

Fig. XI-6. (a) Photograph of mixer IC and (b) mixer module. (c) Measured constellationand spectra of dual-channel back-to-back data transmission experiment [9].
XI-7. Beamforming in 300-GHz BandThe advantage of the 300-GHz band is that it enables the use of a wide frequency range.
However, large free-space path loss occurs during signal propagation through space.
Beamforming technology using a phased-array configuration is an effective solution toextend signal coverage. High-density packaging technology is required for the 300-GHz
band because the chip dimensions (1–1.5 mm) are larger than the half-wavelength
required for beamforming. An example of a successful implementation is a 4-element
300-GHz-band phased-array utilizing a unique flip-chip-on-chip technique that combines
both Si-based and InP-based ICs [10]. The proposed 4-element phased-array transmitterconsists of four CMOS chips, four InP HBT chips, and a PCB substrate (Fig. XI-7(a)).
The CMOS chips were fabricated in a 65-nm CMOS process and include the IF
amplification, LO chain, and mixer. The power amplifier chips were fabricated in an InP
HBT process. The RF frontend on the fabricated module is small. The phased-arraytransmitter achieves a maximum EIRP of 8.4 dBm and has a –6 dB beam coverage of36°, as well as a maximum data rate of 30 Gbit/s over a 50-cm link distance as shown in
Fig. XI-7(b).

69Fig. XI-7. (a) Proposed implementation technique and (b) photograph of measurement
system for EIRP, beam pattern and data transmission [10].
XI-8. Conclusion
This chapter presented the research activities for sub-THz band device technologies
from high power transistors to RF components. These research activities focus on the
research target for sub-THz band mobile communications, which identifies to achieve
the throughput of 100 Gbit/s and the distance from base station to terminal of 100 m.
The research results show excellent performance of each research activity at sub-THz
band. However, the research activities are basic stage for evaluating various
performance. There are some technical issues for realizing the devices for operating at
sub-THz band. In the future, sub-THz band mobile communications will be provided by
the research and development results from the research activities.
XI-9. Acknowledgements
This chapter includes a part of results of “R&D for Expansion of Radio Wave Resources(JPJ000254),” commissioned by The Ministry of Internal Affairs and Communications,Japan.
REFERENCE[1]NTT

DOCOMO,“WhitePaper

5G

Evolution

and

6G”,

https://www.docomo.ne.jp/english/corporate/technology/whitepaper_6g/[2] Hiroto Yamamoto, Atsushi Fukuda, Sumire Aoki, Hiroshi Hamada, Hiroshi Okazaki,
and Yasunori Suzuki, “A Wideband Single Carrier Transmitter Configuration
Equalizing Non-flat Frequency Response,” in The 27th Asia-Pacific Conference on
Communications (APCC 2022), Oct. 2022.
[3] Sumire Aoki, Hiroshi Hamada, Hiroto Yamamoto, Atsushi Fukuda, Teruo Jyo,
Hiroyuki Takahashi, Hiroshi Okazaki, and Yasunori Suzuki, “A Design of InP-HBT

70

100-GHz Diode Linearizer Toward 6G,” in The 27th Asia-Pacific Conference on
Communications (APCC 2022), Oct. 2022.
[4] N. Kashio et al., “Over 450-GHz ft and fmax InP/InGaAs DHBTs With a Passivation
Ledge Fabricated by Utilizing SiN/SiO2 Sidewall Spacers,” IEEE Transactions on
Electron Devices, vol. 61, no. 10, pp. 3423–3428, Oct 2014.
[5] Yusuke Kumazaki, Shiro Ozaki, Naoya Okamoto, Naoki Hara, and Toshihiro Ohki,“Low-Resistance and Low-Thermal-Budget Ohmic Contact by Introducing Periodic
Microstructures for AlGaN/AlN/GaN HEMTs,” IEEE Transactions on Electron
Devices, vol. 69, no. 6, pp. 3073 – 3078, Jun. 2022, doi: 10.1109/TED.2022.3169728[6] Shiro Ozaki, Yusuke Kumazaki, Naoya Okamoto, Naoki Hara, and Toshihiro Ohki,“Improved fT/fmax in wide bias range by steam-annealed ultrathin-Al2O3 gate
dielectrics for InP-based high-electron-mobility transistors,” Applied Physics Express,
vol. 15, p. 041001, doi: 10.35848/1882-0786/ac5a17[7] Yusuke Kumazaki, Shiro Ozaki, Naoya Okamoto, Naoki Hara, Yasuhiro Nakasha,
Masaru Sato, and Toshihiro Ohki, “High-Efficiency 250-320 GHz Power Amplifiers
Using InP-Based Metal-Oxide-Semiconductor High-Electron-Mobility Transistors,”
IEICE Transactions on Electronics, vol. E106-C, no. 11, pp. 661 – 668, Nov. 2023, doi:
10.1587/transele.2023MMP0005[8] V. Petrov, T. Kürner, and I. Hosako, "IEEE802.15.3d: First Standardization Efforts
for Sub-Terahertz Band Communications toward 6G,” IEEE Communications
Magazine Nov. 2020, pp. 28-33.
[9] T. Jyo, H. Hamada, T. Tsutsumi, I. Abdo, S. Kawahara, D. Kitayama, M. Nagatani,and H. Takahashi, “220-to-320-GHz Fundamental Mixer in 60-nm InP HEMTTechnology Achieving 240-Gbps Dual-Band Data Transmission,” IEEE Transactions
on Microwave Theory and Techniques, early access, Nov. 3, 2023, doi:
10.1109/TMTT.2023.3327478.
[10] I. Abdo, T. Jyo, A. Pander, H. Wakita, Y. Shiratori, M. Muto, H. Hamada, M.
Nagatani, C. da Gomez, C. Wang, K. Hatano, C. Liu, A. A. Fadila, J. Pang, A. Shirane,
K. Okada, and H. Takahashi, “300-GHz-Band Four-Element CMOS-InP Hybrid
Phased-Array Transmitter With 36circ Steering Range,” IEEE Microwave andWireless Technology Letters, Vol. 33, Issue 6, Jun. 2023, pp. 887 - 890

71

XII. 6G Simulator using Real Environment Model based on Point Cloud
Data
Koshiro Kitao, Satoshi Suyama, Takahiro Tomie, Nobuaki Kuno,
Kiichi Tateishi, Huiling Jiang
NTT DOCOMO, INC.

Abstract— When studying the deployment of a 5G or 6G system in a specificenvironment such as an indoor environment or factory, it is important to evaluate system
performance beforehand such as throughput in the target environment. In addition, theability to visualize system performance can provide very useful information in deciding
on a method for deploying the communications system. We have developed a 6G systemlevel simulator (6G simulator) to clarify the feasibility of 100 Gbps ultrafast
communications by utilizing the sub-terahertz band as a system, however, existing 6G
simulators can evaluate throughput when using the 100 GHz band in prepared scenarios
such as shopping malls, but not in specific real environments such as indoors or factories .
With the aim of evaluating system performance in real environments, we have developed
an enhanced simulator that can evaluate and visualize 5G and 6G throughput based on
propagation characteristics obtained by ray tracing calculations that employ a realenvironment model based on point cloud data. In this white paper, we describe an
overview of the functions of this simulator and an example of performance evaluation.
XII-1. Introduction
When studying the deployment of a 5G or 6G system in a specific environment such as
an indoor environment or factory, it is important to evaluate system performance
beforehand such as throughput in the target environment. In addition, the ability tovisualize system performance can provide very useful information in deciding on amethod for deploying the communications system. We have developed a 6G system level
simulator (6G simulator) to clarify the feasibility of 100 Gbps ultrafast communications
by utilizing the sub-terahertz band as a system, however, existing 6G simulators canevaluate throughput when using the 100 GHz band in prepared scenarios such as
shopping malls, but not in specific real environments such as indoors or factories [1].
Estimating throughput with high accuracy requires accurate estimation of the radio
propagation characteristics in the environment where the communications system is tobe deployed. A technique for estimating propagation characteristics with high accuracy
that has been attracting attention in recent years is to use point cloud data obtained
from the evaluation environment to create a polygon model of structures and to use thatmodel for ray tracing calculations [2]-[5]. With the aim of evaluating system performancein real environments, we have developed an enhanced simulator that can evaluate and72visualize 5G and 6G throughput based on propagation characteristics obtained by ray
tracing calculations that employ a real-environment model based on point cloud data [6].
In this white paper, we describe an overview of the functions of this simulator and an example
of performance evaluation.

XII-2. Overview of 6G Simulator Based on a Real Environment Model Using Point
Cloud
We have developed a “6G simulator based on a real environment model” with the aimof evaluating throughput in a real environment. This simulator can evaluate andvisualize throughput in 5G and 6G based on propagation characteristics obtained by ray
tracing calculations using point cloud data. Though based on the 6G simulator reported
in [1], this simulator uses information on the propagation characteristics obtained by ray
tracing calculations as a channel model between BSs and MSs used in system-level
simulations. Specifically, the information used by this simulator consists of the
propagation loss, angle of arrival of arriving waves, and propagation delay calculated by
ray tracing.
The evaluation environment used in this simulation is shown in Fig. XII-1. This
simulator can display an image of any environment targeted for evaluation using thepoint cloud data obtained for that environment. The figure shows the image generatedusing the point cloud data obtained in a conference room. This image also reproduces theactual conference room in color by using camera-based image data obtained at the sametime as point cloud data. The simulator can place BSs and MSs at any positions, andhere, the figure shows an evaluation environment for two BSs and six MSs. Whenmultiple BSs are deployed in the simulation, the throughput of an MS at any position
can be evaluated by inputting into a tool the ray-tracing calculation results when MSs
are deployed within the evaluation area for each BS. However, MS movement is notsupported here.

73

BS
MS

Fig. XII-1. 6G simulator evaluation environment using point cloud data
This simulator can also display color maps for several propagation parameters so thatthe relationship between propagation characteristics and throughput characteristics canbe visually grasped. Ray tracing calculation specifications are listed in Table XII-1. Inthese calculations, the polygon data generated by point cloud data for the conference
room shown in Fig. XII-1 is input into Wireless Insight, a commercially available ray
tracing tool. The center frequencies used here are 4.7, 28, and 100 GHz assuming 5G and
6G. The type of antenna used by both BSs and MSs is an onmi-directional antenna, BS
and MS antenna heights are 2.0 m and 1.5 m, respectively, and the condition for ray
searching is 7 reflections. The calculations are performed assuming that the material ofwall surfaces is concrete.
Table XII-1. Ray tracing specifications
項目

諸元

Ray tracing tool

Wireless InsightCenter frequency

4.7 GHz，28 GHz，100 GHz

BS and MS antenna height

1.5 m

BS and MS antennas

Omni-directional antenna (0 dBi)

Ray launching interval

0.25 °

Number of reflections

7

Material

Concrete74

XII-3. Simulation Results
Examples of propagation characteristics calculated by ray tracing are shown in Fig.
XII-2. This figure consists of color maps showing the received level at 100 GHz and theangle spread on the MS side in the horizontal plane. From these maps, it can be seenthat the received level is high near each BS and that the angle spread becomes larger
near the walls of the conference room due to the effects of reflection.
Next, we describe throughput characteristics calculated by this simulator based on the
propagation parameters calculated by ray tracing as described above. The specifications
for system-level simulation of this simulator are listed in Table XII-2. The simulator
generates a fading channel based on each ray’s propagation loss, propagation delay, andarriving angle calculated by ray tracing and calculates throughput when performing BF
by massive MIMO and MIMO spatial multiplexing. Here, user throughput on the DL at
100 GHz is shown in Fig. XII-3. The figure shows user throughput for MS #0–5 circled
in red. It can be seen that MS #3 and MS #4 located in the areas of Fig. XII-2 having ahigh received level and large angle spread can achieve high throughput.
In a similar manner, this simulator could also display user throughput on the DL at
4.7 and 28 GHz. The throughput average of six MSs at 4.7, 28, and 100 GHz was foundto be 0.62, 1.9, and 37 Gbps, respectively. These results confirm that throughput can be
improved by using high frequency bands to provide a wider bandwidth.

Received level map

-60

-50

Angle spread map in horizontal plane

-40

-30

-20

-10

0

0

Received level (dBm)

30

60

90

Angle spread (deg.)

Fig. XII-2. Received level map at 100 GHz and angle spread map on MS side inhorizontal plane

75

Table XII-2. Transmission simulation specifications
Center frequency

4.7 GHz28 GHz

100 GHz

Bandwidth

100 MHz

400 MHz

8.0 GHz

Total transmission powerin TRP

30 dBmMS transmission power

23 dBmNumber of antenna
elements in TRP(vertical × horizontal ×
sub-arrays)

144(4 x 4 x 9)2304
(16 x 16 x 9)

Number of antenna
elements in MS(vertical × horizontal ×
sub-arrays)

144(4 x 4 x 9)

Number of TRPs

2

Number of MS

6

Number of MIMO layers

1, 2, 3, 4, 8

MS#4

MS#5

71.8 Gbps

12.4 Gbps
MS#3

MS#2

9216
(32 x 32 x 9)

84.5 Gbps

MS#1

21.1 Gbps

MS#0

21.1 Gbps

10.6 Gbps

Fig. XII-3. User throughput on DL at 100 GHz
XII-4. Conclusion
We describe an overview of the functions of 6G simulator based on point cloud data
and an example of performance evaluation. Going forward, we plan to compare
throughput measured using experimental equipment in a real environment with thecalculation results of this simulator to evaluate simulator performance and improve it
accuracy. Additionally, with an eye to developing tools that can dynamically control andoptimize 6G through CPS, we plan to conduct technical studies on raising the accuracy

76

and speed of propagation simulations using real environment models and transmission
simulations at the link and system levels.
REFERENCE[1] T. Okuyama et al.: “6G System-level Simulator—Toward 100 GHz Band, 100 Gbps
Extreme-high-data-rate Communications—,” NTT DOCOMO Technical Journal, Vol.
23, No. 3, pp. 13–26, Jan. 2022.
[2] J. Jarvelainen, K. Haneda, and A. Karttunen: “Indoor Propagation Channel
Simulations at 60 GHz Using Point Cloud Data,” IEEE Transactions on Antennas
and Propagation, Vol. 64, No. 10, pp. 4457–4467, Oct. 2016.
[3] P. Koivumaki, G. Steinbock, and K. Haneda: “Impacts of Point Cloud Modeling on the
Accuracy of Ray-Based Multipath Propagation Simulations,” IEEE Transactions on
Antennas and Propagation, Vol. 69, No. 8, pp. 4737–4747, Aug. 2021.
[4] W. Okamura, R. Lukita, G. Ching, Y. Matsuyama, Y. Kishiki, Z. Chen, K. Saito, and
J. Takada: “Simplification method of 3D point cloud data for ray trace simulation inindoor environment,” IEICE Communications Express, Vol. 9, No. 6, pp. 182–187,
Jun. 2020.
[5] K. Kitao, M. Nakamura, T. Tomie, and S. Suyama: “Study of Raytracing using Point
Cloud Data for Indoor Area Evaluation,” ICETC 2022, S1-6, Nov. 2022.
[6] K. Tateishi et al.: “Enhancement of 6G System-level Simulators,” NTT DOCOMOTechnical Journal, Vol. 25, No. 2, Oct. 2023.

77

XIII. THz Sensing and Communication Towards Future Intelligence
Connected Networks
Liu Qiao, Huawei Technologies
Koshimizu Takashi, Huawei Technologies Japan
XIII-1. Introduction
It is envisioned that B5G and 6G would have intelligent connection of everything [1].
Specifically, the demanding ISAC feature would be a novel key service besides the
exploration of the THz band. We have key technology research and prototype verification,
which shows 210 Gbit/s data transmission rate at 330 m distance, and up to 3 mm
invisible imaging, which is the highest performance in ISAC field.
XIII-2. THz Spectrum and Application Scenarios
A number of bands have been recently approved for ultra-high data rate transmission,
which motivates research on THz spectrum and application scenarios.
XIII-3. THz Channel Propagation and Modeling
A hybrid channel modeling methodology is proposed. A few suggested methods are listin Table XIII-1.
Table XIII-1. Terahertz channel modeling methodology
Application
Channel Modeling Method
Communication
Stochastic
Positioning

Stochastic (GBSCM)

Localization and mapping
Imaging and recognition

Ray-based
EM-based

XIII-3.1. Channel Measurement System and Measurement Campaign
The measurements are achieved at 140, 220, and 280 GHz. Parameters are listed inTable XIII-2.
Table XIII-2. Parameters of the measurement system
Parameter
Value
Frequency band [GHz]140
220
Local oscillator [GHz]10.66718
Start frequency [GHz]130
201
End frequency [GHz]143
209
Bandwidth [GHz]13
8
Sweeping points
1301
801
Transmitter antenna gain [dBi]15
10Receiver antenna gain [dBi]
25
25

78

Azimuth rotation range [degree]
Elevation rotation range [degree]
Delay resolution [ps]Maximum excess delay [ns][0:10:360][–20:10:20]
76.9
100

[0:10:360][–20:10:20]
125
100

XIII-3.2. THz Channel Characterization and Analysis
The multi-frequency alpha-beta-gamma (ABG) path loss results on 140, 220, and
280 GHz bands are depicted in Fig. XIII-1. The proposed ABG path loss model is asfollows:
𝑑

𝑓

0

0

𝑃𝐿𝐴𝐵𝐺 [𝑑𝐵] = 20.7𝑙𝑜𝑔10 (𝑑 ) + 26.72 + 22.2𝑙𝑜𝑔10 (𝑓 ) + 2.53

(2)More measurement campaign and modeling results can be referred to that in theliterature [2–3].
(a) Delay-angular spread

Fig. XIII-1. ABG path loss model results at the meeting room
On the small-scale aspect, the THz channel also exhibits different propagation. For
further study, we choose one receiver position, as shown in Fig. XIII-2 to analyze thespatial angle of arrival (AoA).

Fig. XIII-2. Indoor hotspot cell (InH) meeting room 140 GHz measurement scenario
XIII-4. THz Hardware and Components
XIII-4.1. THz Components
Fig. XIII-3 demonstrates a GaN HEMT power amplifier and its package operation
at 220 GHz. The saturation output power reaches 18 dBm. The best amplification
application for CMOS is at 140 GHz, and the gain can be achieved at 200 to 300 GHz

79using positive feedback technology [6]. BiCMOS currently operates at a maximum
amplification frequency of 310 GHz, achieving 4 dBm of output power [5].

Fig. XIII-3. Output performance of 220 GHz GaN HEMT power amplifier
THz receivers can be classified as heterodyne and direct detector receivers. The
different technologies available to build THz and sub-THz receivers are shown in Fig.
XIII-4.

Fig. XIII-4. THz receivers with respect to source technology, with 1 Hz resolution
bandwidth (RBW)
XIII-4.2. THz Antenna
In a THz "tile-able" array, large number of antennas are integrated on chip. High-level
integration and scalability are two important considerations in tile design.
Programmability is another desired feature of THz "tile-able" array. Limited
configurability has been demonstrated mainly through electrically, mechanically or
thermally controlled reconfigurable materials [8–9]. In addition, CMOS integration is an
important consideration to allow for low-cost fabrication.
XIII-4.3. Intelligent Surface
In [10], intelligent surface is classified into passive surface and active surface. Fig.
XIII-5 shows a wireless communication environment enabled by intelligent surfaces.

80(a) Outdoor environment(b) Indoor environment

Fig. XIII-5. Typical RIS use cases for THz
XIII-4.4. Integration Technologies
Multichip module (MCM), system-in-package (SiP), and heterogeneous integration
are promising candidates. Antenna and silicon lenses are integrated in the package toreduce connection loss and enhance EIRP [11]. The through-silicon via (TSV) process
has better integration and process precision and can be used at higher frequencies [12].
The embedded wafer level ball grid array (eWLB) technology usually used at low
frequency can also be used in lower end of THz [13] as SiP technology. Wafer-level
integration using Benzo cyclobutene (BCB) provides a 2D integration method [14]. The
alternative approach bonds the wafer or die of III-V materials onto a patterned siliconwafer [15].
XIII-5. THz System and Testing
XIII-5.1. THz Communication System
Our THz communication platform at 220 GHz frequency, in which multiple
advanced technologies are realized. Field trial experiment was conducted as shown in
Fig. XIII-6 [16].

Fig. XIII-6. THz field trial, 2 x 2 polarization-MIMO system

81

Maximum throughput is 210 Gbit/s, and net data rate is 168 Gbit/s. The demodulated
constellation is shown in Fig. XIII-7.

Fig. XIII-7. 64QAM demodulated constellation
Long distance field trial for single-input single-output (SISO) is also considered in our
verification. As shown in Fig. XIII-8, the link distance is 3.6 km with high-humidity
weather.

Fig. XIII-8. 3.6 km long distance test
Maximum throughput is 35 Gbit/s, and net data rate is 28 Gbit/s after removal of the
FEC overhead.

Fig. XIII-9. 3.6 km, demodulation constellation
To explore THz-ISAC, we use a similar system architecture and device. By using theconcept of virtual MIMO and compression sensing algorithms, the EM imaging of metal
objects hidden in a paper box is rebuilt successfully, and mm-level imaging resolution isachieved, as presented in Fig. XIII-10.

Fig. XIII-10. Imaging results of non-sparse full aperture scanning

82

XIII-5.2. THz Sensing System
We developed both simulation model and measurement setup for THz-TDS material
characterization application. As shown in Fig. XIII-11, both permittivity and loss
tangent agree well with the theoretical values. [17] showed a quasi-optical system that
performs complex material property measurement. The complex permittivity ofvarious Rogers RT/duroid series PCB substrates agreed well with the literature.

Fig. XIII-11. Simulated and measurement results of material characterization
application; Measurement from [60].
In addition, a qualitative microwave holography (QMH) imaging method wasdemonstrated to perform the imaging and material mapping application [18–19]. The
image test-bed was then set up as shown in Fig. XIII-12. It is shown that QMH canachieve spatial resolution close to λ/4, even under a far field measurement set up. A
comparison can be found in [19–20].

Fig. XIII-12. QMH reconstruction of metal and nylon balls of various sizes. Resultsfrom [20]83XIII-6. Conclusion
In this paper, we discussed the advantages and typical scenarios of THz
communication and sensing applications. We also proposed a hybrid channel modelingframework to improve the modeling accuracy and efficiency. In particular, the THz
subsystem with silicon and III-V compound semiconductor materials heterogeneous
integration is analyzed and proposed to improve the performance by using the advantage
of different processes and materials. Finally, the prototype and measurement campaigns
were conducted to illustrate the advantages for high throughput communication andhigh-resolution sensing scenarios. A variety of measurement campaign examples show
210 Gbit/s data transmission rate at 330 m distance, and up to 3 mm invisible imaging,which is the highest performance in this field.

References[1] W. Tong and P. Zhu, "6G: the next horizon, from connected people and things toconnected intelligence," Cambridge University Press, 2021.
[2] He J, Chen Y, Wang Y, et al., "Channel measurement and path-loss characterization
for low-terahertz indoor scenarios," arXiv preprint arXiv:2104.00347, 2021.
[3] Y. Chen, C. Han, Z. Yu, and G. Wang, "140 GHz channel measurement andcharacterization in an office room," ICC 2021 - IEEE International Conference on
Communications, 2021, pp. 1-6, doi: 10.1109/ICC42927.2021.9500596.
[4] Y. Chen, Y. Li, C. Han, Z. Yu, and G. Wang, "Channel measurement and ray-tracingstatisticalhybridmodelingforlow-terahertz

indoorcommunications,"

arXiv:2101.12436[5] U. R. Pfeiffer, R. Jain, J. Grzyb, S. Malz, P. Hillger, and P. Rodríguez-Vízquez,"Current status of terahertz integrated circuits - from components to systems," 2018
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[6] X. Li et al., "A 250-310 GHz power amplifier with 15-dB peak gain in 130-nm SiGe
BiCMOS process for terahertz wireless system," in IEEE Transactions on Terahertz
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no. 5, pp. 118-125, October 2020, doi: 10.1109/MWC.001.1900534.
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26, no. 4, pp. 291-293, April 2016, doi: 10.1109/LMWC.2016.2537044.
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Technology, vol. 1, no. 2, pp. 260-267, Feb. 2011, doi: 10.1109/TCPMT.2010.2099731.
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1414, Jul. 2019.
[14] X. Yang et al., "Low-loss heterogeneous integrations with high output power radar
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11, no. 1, pp. 2-15, Jan. 2021, doi: 10.1109/TTHZ.2020.3036181.
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85

XIV. 300GHz point-to-point fixed wireless transmission systems
Tetsuya Kawanishi, Waseda University
Riku Yoshino, Waseda University
Arata Ogaki, Waseda University
Kunihisa Jitsuno, Waseda University

Abstract—In future mobile networks, many small cells will be needed to provide
broadband service everywhere. In order to efficiently connect many base stations, mobile
backhaul and fronthaul should be constructed by various types of transmission media,
including short-range fixed radio transmission in the THz band. This paper describes a
300 GHz fixed radio transmission system based on V-band IF signal processing.
IEEE802.15.3e compliant 60 GHz transceivers are used as baseband and IF units. To
expand the flexibility of the network configuration, THz fixed radio can use non-line-ofsight propagation due to reflection as in conventional microwave systems. However, theeffect of reflector roughness must be considered. This paper also discusses recent
research results on reflection loss and scattering due to roughness.
XIV-1. Introduction
Many small cells would be required to provide broadband services anywhere in future
mobile networks. To connect many base stations effectively, mobile backhaul andfronthaul should be constructed by various types of transmission media including THz
short distance fixed wireless transmission. Optical fiber links are commonly used formobile backhaul and fronthaul in north-east Asia, while fixed wireless links are often
used in developing countries [1]. To mitigate huge traffic demand in mobile networks,
conventional fixed wireless links using microwave bands would be replaced by opticalfibers. However, it would be rather difficult to make all the base stations connected only
by optical fibers, because the number of the base stations would be huge. A part of mobile
backhaul or fronthaul links would be by fixed wireless connections, where fiber
installation cost is high [2]. The fixed wireless and optical transmission offer
heterogeneous networks where high-speed optical fiber links are bridged by high-speed
fixed wireless links. The required transmission capacity is much higher than that ofconventional fixed wireless transmission using microwave. On the other hand, therequired transmission distance is much shorter than a few hundred meters. Such fixedwireless links should be based on millimeter-wave or THz radio, where large bandwidths
are available for fixed wireless transmission and attenuation in the air is relatively large.
Normally, fixed wireless links are based on line-of-sight (LoS) propagation, however,
non-line-of-sight (NLoS) propagation with reflectors also can be used to enable86installation of fixed wireless links in various locations. Fixed reflectors or passive
repeaters have been used in conventional microwave fixed wireless links to increasetransmission range [3]. Reflectors can be also used in millimeter-wave or THz bands.
However, the reflection loss would be larger than in microwave bands, because ofroughness. Typical roughness of smooth wooden surfaces is about 100 micrometers.
In this paper, we will describe an overview of millimeter-wave and THz fixedtransmission technologies where performance degradation due to weather conditions
should be taken into account [5-8]. For long term evaluation, we are developing a 300GHz
transmission system which can record data of wireless transmission and that of weather
conditions. This paper also reviews recent research activities on THz-wave scatteringfrom slightly rough surfaces to investigate reflection loss due to roughness [9, 10].
XIV-2. 300GHz Fixed wireless transmission using V-band modems
Fig. XIV-1 shows a schematic of a networks connecting baseband units and remoteantenna units through mobile backhaul and fronthaul links consisting of wired andwireless transmission. Optical fibers are commonly used for high-speed wired
transmission. To avoid the wireless link becoming a bottleneck, the transmission
capacity of the wireless link connecting to the optical fiber must be comparable to thatof the optical fiber transmission. On the other hand, the required transmission range forthe wireless link would be less than one kilometer, because separation between the
remote antennas would be less than a few hundred meters. Many antennas are for small
cells in future mobile networks. Fixed wireless systems using millimeter or THz bands
can offer high-speed and short-distance radio links.
Over 100Gb/s wireless transmission has been recently demonstrated by using
quadrature amplitude modulation with polarization multiplexing [11, 12]. However,most of research results are based on offline signal processing or unidirectionaltransmission demonstration. Duplex wireless systems are required to provide data links
and functions over TCP/IP. MIMO also needs duplex links. Recently, 300 GHz duplex
high-speed transmission was demonstrated by using millimeter-wave modems as
baseband and IF modules [13]. V-band or E-band millimeter-wave signals are converted
to 300GHz signals by using broadband mixers.

87Fig. XIV-1. Mobile backhaul and fronthaul with fixed wireless links [2]Fig. XIV-2 shows a configuration of the 300GHz transmission system using E-band
FDD modems. E-band signals from four modems are combined together by a millimeterwave circuit and upconverted into a 300GHz signal. A TDD system was also
demonstrated by using V-band modems which comply with IEEE 802.15.3e. Two
802.15.3e signals can be aggregated and upconverted into a 300GHz TDD signal, as
shown in Fig. XIV-3. Based on this research result, outdoor radio units with weather
condition logger are being developed for long-term evaluation. 300GHz power amplifiers
and mixers are connected to two V-band modems for parallel TDD duplex transmission.
All the components are assembled in a water-proof box as shown in Fig. XIV-4. An
antenna is shared with transmission and reception through a coupler with band-bassfilter. IF signals in two 802.15.3e channels whose center frequencies are 60.48GHz and62.64GHz are upconverted into 286.20GHz and 288.36GHz which are corresponding tochannels 16 and 17 of IEEE 802.15.3d. 300GHz output power from an output port of theoutdoor unit is about 0dBm and the antenna gain is 40dBi. As shown in Fig. XIV-5, apair of outdoor radio units are installed in motor-controlled platforms for automaticalignment. The platform can control angle in 360 degrees in azimuth and from -45
degrees to +20 degrees in elevation with 0.01-degree resolution. The automaticalignment function would facilitate installation process and offer adaptable network
configuration. We have confirmed that the outdoor units can offer stable wireless datatransmission whose bitrate is 4Gb/s with QPSK. The THz signal complies with IEEE
802.15.3d.

88

Fig. XIV-2. Configuration of 300GHz transmission system using E-band modems [14]Fig. XIV-3. Configuration of 300GHz transmission system using V-band modems [14]Fig. XIV-4. 300GHz outdoor radio unit89Fig. XIV-5. Outdoor radio unit mounted on a motor-controlled platform.
XIV-3. THz-wave scattering
THz-wave scattering from slightly rough surfaces can be simulated by using thestochastic functional approach, where the scattered wave can be expressed as afunctional of rough surfaces [9, 10]. The rough surface is expressed by two parameters:
surface roughness, , and correlation length, l. To focus on the scalability of thescattering problem, normalized surface roughness, k  and correlation length, kl, arecommonly used to describe the surfaces. Typical roughness of smooth wooden surfaces
would be 40-100um, which corresponds to k  of 0.2-0.7. Fig. XIV-6 shows examples ofcalculated scattering wave distribution which is normalized by the incident wave power[15]. The left graph shows horizontal (TE) polarization. The right shows vertical (TM)polarization. The surface parameters for 300GHz, k , kl, and n are respectively 0.3074,
0.419 and 1.3, which are actual values of a wooden sample. The roughness andcorrelation length are measured by a digital microscope.
The loss in reflection due to scattering can be calculated from integration of scattering
distributions over a celestial sphere. The total scattering wave power is subtracted fromthe reflected and refracted wave power on a flat surface. Fig. XIV-7 shows the loss forvarious roughness. The loss is an increasing function of roughness k . When theroughness is larger than 0.5, the loss would not be negligible.

90Fig. XIV-6. Scattering wave distribution calculated by stochastic functional approach,(left) horizontal polarization incidence, (right) vertical polarization incidence.

Fig. XIV-7. Reflection loss due to roughness.
XIV-4. Conclusion
In this paper, we presented millimeter-wave and THz fixed transmission technologies
which can be used in mobile backhaul and fronthaul in future mobile networks. We aredeveloping 300GHz outdoor radio units which can record weather conditions andtransmission performance for long term evaluation. NLoS propagation with reflectors
also can be used to enable installation of fixed wireless links in various locations.
However, the reflectivity would be degraded by scattering on rough surfaces. This paper
reviewed recent research activities on THz-wave scattering from slightly rough surfaces
to investigate reflection loss due to roughness.
XIV-5. Acknowledgements
This presentation includes research results were supported by / obtained from ThoR
(TeraHertz end-to-end wireless systems supporting ultra-high data Rate applications)funded in part by the Horizon 2020, the European Union's Framework Programme for
Research and Innovation, under Agreement 814523, and in part by the Commissioned

91

Research

(No.

196),

JPJ012368C00302,

JPJ012368C00401,

JPJ012368C43103,

JPJ012368C04901, JPJ012368C06001. THz transmission study was partly conductedunder the contract “R&D of high-speed THz communication based on radio and opticaldirect conversion” by the Ministry of Internal Affairs and Communications of Japan
under Grant JPJ000254.
REFERENCE[1] Mobile backhaul options, Spectrum analysis and recommendations, GSMA, 2018https://www.gsma.com/spectrum/wp-content/uploads/2019/04/Mobile-BackhaulOptions.pdf[2] T. Kawanishi, Wired and Wireless Seamless Access Systems for Public Infrastructure,
Artech House 2020.
[3] APT Report on Technologies of Fixed Wireless System to Provide Remote
Connectivity, APT/AWG/REP-85 (09/2018)[4] E. Csanady, et al.: Quality of Machined Wood Surface (Springer, 183, 2015) (DOI:
10.1007/978-3-319-22419-0_4).
[5] Z. -K. Weng et al., Millimeter-Wave and Terahertz Fixed Wireless Link Budget
Evaluation for Extreme Weather Conditions, IEEE Access, 9, 163476-163491 (2021)[6] U. Mankong et al., Millimeter Wave Attenuation Due to Wind and Heavy Rain in a
Tropical Region. Sensors, 23, 2532 (2023)[7] APT Recommendation on Models for FWS Link Performance Degradation due to Wind,
APT/AWG/REC-12 (12/2022)[8] APT Report on FWS link performance under severe weather conditions,
APT/AWG/REP-81 (04/2018)[9] R. Yoshino et al., Modeling and evaluation of millimeter wave scattering fromminimally rough surfaces on stones, IEICE Electronics Express, 19, 20220257 (2022)[10] R. Yoshino et al., 300GHz Terahertz Wave Scattering Experiment and Simulationfrom Slightly Rough Surfaces on Dielectrics (to be published in IEICE

Communications Express)[11] W. Tong et al., Experimental Comparison of Single-Carrier and Digital Subcarrier
Multiplexing Transmissions in a W-Band 200 Gb/s Fiber-Wireless System
Considering Transmitter IQ Imbalance and Skew Mitigation, IEEE Photonics Journal,
15, 1-8 (2023)[12] T. Kawanishi, THz and Photonic Seamless Communications," in Journal ofLightwave Technology, 37, 1671-1679 (2019)[13] ThoR: TeraHertz end-to-end wireless systems supporting ultra high data Rateapplications, https://thorproject.eu/

92

[14]

ThoR

Deliverables

D2.2

Overall

SystemDesignhttps://thorproject.eu/wp-

content/uploads/2019/07/ThoR_SIKLU_190417_F_WP2-D2.2-Overall-SystemDesign.pdf[15] R. Yoshino et al., Scattering experiment and the effect of polarization from wooden
surface for terahertz wave, IEICE Tech. Rep., vol. 123, no. 351, MWPTHz2023-86, pp.
60-63, Jan. 2024.

93Abbreviation List
Abbreviation

Explanation

ABG

Alpha-Beta-Gamma

AI

Artificial Intelligence

ALD

Atomic Layer Deposited

AMC

Adaptive Modulation and Coding

AoA

Angle of Arrival

AR

Augmented Reality

ASIC

Application Specific Integrated Circuit

AWG

Arbitrary Waveform Generator

BAN

Body Area Network

BCB

Benzo cyclobutene

BER

Bit Error Rate

BF

Beam Forming

BS

Base Station

CC

Component Carrier

CI

Close-in

CMOS

Complementary Metal Oxide Semiconductor

CPS

Cyber Physical System

CSI

Channel State Information

DC

Direct Current

DFT

Discrete Fourier Transform

DL

Down Link

DNN

Deep Neural Network

DOA

Direction of Arrival

DSP

Digital Signal Processing

EVM

Error Vector Magnitude

eWLB

embedded Wafer Level Ball grid array

FDD

Frequency Division Duplex

FDE

Frequency Domain Equalize

FSPL

Free Space Path Loss

HARQ

Hybrid Automatic Repeat Request94Abbreviation

Explanation

HPBW

Half Power Beam Width

IBO

Input Back Off

IFFT

Inverse Fast Fourier Transform

InH

Indoor hotspot cell

ISAC

Integrated Sensing and Communication

ITU-R

International
Telecommunication
Radiocommunication Sector
Key Performance Indicator

KPI
LAN

Local Area Network

LNA

Low-Noise Amplifier

LOS

Light of Sight

LTE

Long Tern Evolution

MCM

Multichip Module

MIMO

Multiple-Input and Multiple-Output

MMIC

Monolithic Microwave IC

MS

Mobile Station

MOS

Metal Oxide Semiconductor

MOS-HEMTMSL

Metal-Oxide-Semiconductor
Transistor
Microstrip Line

Eigh-Electron-Mobility

NLOS

Non-Line of Sight

NR

New Radio

NRNT

New Radio Network Topology

OAM

Orbital Angular Momentum

OFDM

Orthogonal Frequency Division Multiplexing

PA

Power Amplifier

PCB

Printed Circuit Board

PLE

Path Loss Exponent

QMH

Qualitative Microwave Holography

RAN

Radio Access Network

RAT

Radio Access Technology

RD

Relay Device

RFRadio Frequency

95

Union

Abbreviation

Explanation

RIS

Reconfigurable Intelligent Surface

RMSE

Root Mean Square Error

RS

Relay Station

Rx

Receiver

SAG

Selective-Area Growth

SC

Single Carrier

SiP

System-in-Package

SISO

Single-Input Single-Output

SIW

Substrate-Integrated Waveguide

SNR

Signal to Noise power Ratio

Sub-THzSub Tera Hertz

TDDTime Division Duplex

TDS

Time Domain Spectroscopy

THz

Tera Hertz

TMA

Trimethylaluminum

TSV

Through-silicon Via

Tx

Transmitter

UCA

Uniform Circular ArrayUE

User Equipment

UL

Up LinkVR

Virtual Reality

96