One-Inductor Single-stage Differential Boost Inverter
Operated in Discontinuous Current Mode forSingle-Phase Grid-Tied Photovoltaic System
Ayato Sagehashi, Le Hoai Nam and Jun-ichi Itoh
Department of Electrical Engineering
Nagaoka University of Technology, NUT
Nagaoka, Niigata, Japansagehashi@stn.nagaokaut.ac.jp, lehoainam@stn.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp
Abstract—In this paper, a novel single-stage boost inverter
using one inductor is proposed for a grid-tied photovoltaic (PV)inverter. The proposed inverter shares one inductor with two
boost converters by utilizing discontinuous current mode (DCM).
The output voltage is controlled by the differential voltage
between boost converters. Using this topology, the neutral pointof the grid can be connected to a terminal of the solar panel. As aresults, common-mode noise can be suppressed. The operation of
the proposed inverter is confirmed with 100-W prototype with a
DC input voltage of 100 V and an AC output voltage of 100 V.
The output current THD is 1.4%.
Keywords—DC/AC converter, Discontinuous current mode,
Micro-inverter

I.

INTRODUCTION

In recent years, PV systems have been widely used as analternative power supply from natural energy. In such system,
the PV inverter is required for interconnecting the grid and the
PV system. In particular, the PV inverter is strongly required tominimize and achieve high efficiency[1]-[5]. The generated
power of the PV system depends heavily on the weather
condition and the solar radiation. Therefore, in theconventional PV systems where only one inverter is connectedwith several solar panels, if only one solar panel is shaded byclouds or rain, the output power of the whole system worsens.
In order to solve this problem, a micro-inverter has been
proposed as an alternative method for PV inverter [6]-[10]. In
particular, each micro-inverters are connected to each solar
panels separately. Therefore, the generated power will be
increased in partial shadow condition because the generated
power of each solar panel is controlled independently.
Consequently, problems on individual solar panels does not
influence heavily on the operation of the whole system.
However, the minimization of the micro-inverter is necessary
in terms of the installation area in order to connect to each solar
panel separately.
One of the main factors preventing the minimization of the
micro-inverter is the passive components, i.e. the inductor andthe capacitor. In general, the PV inverter is composed of aboost converter and an inverter. The boost converter is used tocontrol the DC-link voltage, whereas the inverter converts the
DC power to AC power. In this case, a high-voltage DC-link
capacitor is required with a large capacitance, which absorbs a
power fluctuation caused by the single-phase grid. On the other
hand, the system volume of the DC/AC converter increases
because an boost inductor of a boost converter andinterconnected inductors of an inverter are necessary.
On the other hand, the single-stage boost inverter for the
DC/AC converters have been proposed in order to unify the
boost converter and the inverter [11]-[15]. The single-stage
boost inverter can reduce the number of the switching devices
and the inductors. It is two inductors for the boost operationand four switching devices compose two boost converter
circuits. Therefore, the single-stage boost inverter can be
minimized due to the reduction of circuit elements comparedwith the typical DC/AC topology. In addition, the leak current
caused by the floating capacitance between the solar panel andthe ground is significantly reduced due to the voltage
fluctuation at the neutral point is small between the solar panel
and the grid. However, the single-stage boost inverter still use
two inductors. Consequently, the number of the inductor isnecessary to further reduce in order to minimize the circuit
volume.
In this paper, a novel boost inverter is proposed in order toconvert from DC to AC including the boost operation using
only one small input inductor. First, the conventional two stage
DC/AC converter for the PV system, the conventional singlestage boost inverter and the proposed boost inverter are
explained. Secondly, the operation modes and switchingpatterns of the proposed boost inverter is shown. Third, theinput inductor current control and the output capacitor voltage
control are explained. In addition, the calculation method ofeach duty ratio command values for DCM is established.
Fourth, the operation of the proposed boost inverter bysimulation after the design method of the input inductor at thecritical mode of DCM is confirmed. Next, the losses of theswitching devices and inductors are compared between theconventional two stage DC/AC converter and the proposed
boost inverter. Finally, the operation of the proposed inverter isconfirmed in experiment. As a result, the conversion from DC

to AC is realized by the proposed boost inverter which reduces
the number of inductors compared with the conventional twostage DC/AC converter and the conventional single-stage boost
inverter.
II.

Lg
L

DC/AC CONVERTER TOPOLOGY

A. Conventional two-stage AC/DC converter
Fig. 1 shows the conventional two stage DC/AC converter
topology for the PV inverter. The conventional two stage
DC/AC converter is composed the boost converter and the Hbridge inverter in order to convert from DC to AC. The DC
link voltage requires the higher voltage than the maximum gridvoltage when the inverter is connected to the grid. Therefore,
the large capacitor is necessary to connect between the boost
converter and the inverter when the input voltage is lower than
the grid voltage. Additionally, the boost inductor and theinterconnected inductors prevent in order to minimize thesystem. As a result, a number of passive components,
switching devices and the value of the capacitance are required
to reduce in order to minimize the system of the conventional
two stage DC/AC converter.

Fig. 4 shows the operation mode of the proposed boost
inverter. The operation mode is shown only the output

Lg

Vg

b
Vin

Fig. 1. Conventional two-stage DC/AC converter.
a

b

Vg
L

L
VCa

Cb

Ca

VCb

Vin

B. Conventional single-stage boost inverter
Fig. 2 shows the conventional single-stage differential
boost inverter. Two boost converters compose the conventional
single-stage differential boost inverter. This inverter regulates
the voltage of output side capacitors VCa and VCb to sinusoidal
waveform including the offset voltage higher than the input
voltage. In addition, the output current is controlled bypotential differences of the output capacitors. Therefore,
compared to the conventional two stage DC/AC converter, theconventional single-stage boost inverter requires only four
switches and two inductors. In this case, the total volume of theboost inductors of the single-stage boost inverter is smaller
than that of the conventional two stage DC/AC converter.
However, it is necessary to further reduce the number of theinductors in order to minimize the system volume.
C. Proposed single-phase boost inverter
Fig. 3 shows the circuit diagram of the proposed singlephase boost inverter. In principle, the proposed boost inverter
is combined of two boost converter, requiring a switching
device and two bidirectional switches. Note that if only typicalunidirectional switches are used, the proposed boost inverter
requires more switching devices compared with theconventional single-stage boost inverter. On the other hand, theproposed boost inverter is operated in DCM. Consequently, thenumber of the input inductor is reduced to only one because theinput inductor current is shared for the voltage control of each
output capacitor Ca and Cb. Hence, the converter size can be
reduced. Furthermore, the number of the switching devices ofthe proposed inverter can be reduced to only three if thebidirectional switches can be employed with GaN-basedmonolithic bidirectional switches, which has same on
resistance as the unidirectional switches [16].

aCdc

Fig. 2. Conventional single-stage differential boost inverter.
Sb

b

Vg
Sa

a

L

S

Cb

Ca

VCa

Vin

VCb

Fig. 3. Proposed single-stage differential boost inverter.
L

L

S1

S2

S

Vin

S1

Ca

S

Vin

(a) Mode I

Ca(b) Mode II

L

L

S1

Vin

S2

S2

S

(c) Mode III

S1
Ca

Vin

S2

S

(d) Mode IV

Fig. 4. Operation mode for each capacitor voltage control.

Cacapacitor Ca caused by the operation mode of the output
capacitor Cb similarly. The output capacitor voltages arecontrolled similar to the boost operation and the step-down
operation. The input inductor current is controlled to positive
direction at Mode I and II periods. At Mode III and IV, theinput inductor current is controlled to negative direction inorder to get the desired output capacitor voltage VCa. In thisoperation mode, the boost inverter operate the boost mode
when the output current command value is positive. The
operation is the step-down mode when the other direction of
the output current command value. Furthermore, the sinusoidal
waveform of the output side is realized by the differential
voltage between output capacitor voltages VCa and VCb whichis controlled to inverse phase.
III.

Capacitor voltage control
iout*
+
+
iLa*
VCa_avg*
PI +iCa* VCa
VCa_avg
Vin

DCM
d3
VCa Vind4
VCa
When iLa*≦0
Fig. 5. Control block for output capacitor Ca voltage control.

CONTROL METHOD FOR BOOST INVERTER

d1+d2+d0+d3+d4

Fig. 5 shows the voltage control block for the output
capacitor Ca in the boost inverter. The PI controller controls
only the average voltage of the output capacitor Ca. In addition,
the output current command is added to the inductor current
command value in the minor loop as a feed forward regulationin order to control the output current [17]. The output capacitor
voltage is necessary to be higher than the input voltage in orderto achieve the sinusoidal waveform at the output side of boost
inverter. Hence, the output capacitor voltage command value
must satisfy (1).


VC _ avg 

Vout_ peak2i La *  (iCa  iout *)Sawtooth Carrier

d1+d2+d0+d3
d1+d2+d0d1+d2d1tMode I

Mode II

Mode III Mode IV

Ipeak
Iave

 Vin 



VCa 
Pout
 iCa 
Vin 
Vout _ rmstCurrent flowing to Ca

where VC_avg is the output capacitor voltage command value,
Vout_peak is the maximum amplitude of the grid voltage, Vin is
the input voltage. The input inductor current command value iscalculated after deciding theoutput capacitor voltage
command value by (1). The input inductor current command
value is decided by (2)When iLa*＞0
DCM
d1
VCa
d2
VCa VinV
2 sin(2f outt ) Ca   Vinwhere iCa is the output capacitor command value voltage
control, iout* is the output current command value, VCa is theoutput capacitor voltage, Pout is the output power, Vout_rms is theeffective value of the grid voltage, fout is the frequency of thegrid. Note that the input inductor current command value forthe other output capacitor voltage VCb is omitted due toexplanation by inverting the output current command value.
The duty of the boost inverter is then compared with sawtooth
waveform to generate the switching signal.
Fig. 6 shows the relationship between the input inductor
current and the duty ratio of the boost inverter. Duty ratio
command values d1 and d2 are the period for the control of theoutput capacitor voltage VCa. Duty ratio command values d3
and d4 are the period for the control of the output capacitor
voltage VCb. These duty ratio command values are controlled inorder to share the input inductor current. Furthermore, the duty
ratio command value d0 is the zero current period in order toCurrent flowing from Cb
I’peak

Fig. 6. Waveform of relationship between inductor current andduty ratio.

prevent the interference of the current control between the
output capacitor Ca and Cb.
The duty ratio for the input inductor current control iscalculated by the relationship between the maximum value andthe average value of the input inductor current. In the control of
the output capacitor voltage VCa, the maximum value and theaverage value of the input inductor current is shown by (3) and
(4).




I peak_ Vca Vin
V  Vind1Tsw  Ca
d 2Tsw LL

I ave_ Vca 

I pea k_ Vca2( d1  d 2 ) 





where L is the input inductor, d1 and d2 are the duty ratio in Fig.
6. The relationship between d1 and d2 is calculated by the
maximum value and the average value of the input inductor
current. The duty ratios d1 and d2 are explained by (5) and (6).



d1 

2 L(VCa  Vin )iave _ Ca* 
VinVCaTsw

2 2 Pout L(VCa  Vin )2

Vin Vout_ rmsTsw





Vpeak
VCa
VC_avg

d2 

Vin
d1
VCa  Vin

(6)

VCb

The inductor current is controlled to the DCM by duty ratio.
Similarly, in based on the output capacitor voltage VCb, theduty ratio of the input inductor current d3 and d4 is calculatedby (7) and (8).

Vin+Vout_margin

VintFig. 7. Relationship between input voltage and output capacitor voltage
at inductor design.

d3 

2 LV ini ave _ Cb* (VCb  Vin )VCbTsw

d4 

2 2 Pou t L 
(VCb  Vin )Vou t_ rmsTsw

VCb  Vin
d3
Vin

(8)

As a result, the DCM operation of the input inductor current isachieved. In addition, the input inductor current share each
operation mode in input inductor.
IV.

The boost inverter shares the current of each operation
modes on only one input inductor. Therefore, the dutycommand values of operation modes are satisfied (9).
d1  d2  d3  d4  1

600

(9)

In this case, the condition is the critical mode; i.e. it is theworst case at the DCM operation of the inductor current. Notethat the duty command value of the zero current period d0 is
added to (9) if the zero current is necessary. The conduction
loss of switching devices can be reduced due to the current
ripple of the inductor current is the minimum at the criticalmode of the DMC operation. Hence, the best condition for the
input inductor is the critical mode of DCM operation at the
rated power. The input inductor value of the critical mode isdesigned by (10) which are relationship between dutycommand values from (5) to (8) and the condition of thecritical mode (9).

Pout=200 W

400

Pout=300 W
200
0

CONSIDERATION OF INDUCTOR

A. Design of input inductor for boost inverter
Fig. 7 shows the waveform of the input voltage Vin, output
capacitor voltages VCa and VCb in the boost inverter. In this case,the average output capacitor voltages are decided by (1).
Furthermore, each output capacitor voltages have a margin
Vout_margin in order to prevent the input voltage and output
capacitor voltages from becoming same voltage.

Pout=100 W

800

Input inductor [mH]



0

100
200
Input voltage Vin [V]300

Fig. 8. Designed inductor value of boost inverter at critical mode of
DCM (include 10% margin).

2

L



Vin Vou t_ rmsTsw
2


 
VCa
VCb

2 2 Pou t  V V

V

V
Ca
in
Cb
in





where Vg_rms is the effective value of the grid voltage, Pout is theoutput power, Tsw is the switching period of the boost inverter.
Moreover, the input inductor value have a 10% margincompared with the design value when the prototype circuit of
the boost inverter use the input inductor designing by (10).
Fig.8 show the designing inductor value when the input
voltage is changed at each output power conditions. The design
conditions are 200 V of the grid voltage and 10 kHz of theswitching frequency. In addition, the minimum voltage of theoutput capacitor is higher than the input voltage by 20% of the
maximum value of the grid voltage. The input inductor isincreased when the input voltage is increased. On the other
hand, the input inductor value become small when the output
power become high.
B. Inductor comparison with conventional system
The value of the input inductor for boost inverter is undermH when the input voltage is changed. Therefore, it isconsidered how the input inductor of the boost inverter can be
decreased compared with the conventional two-stage system.

Fig. 9 shows the structure of the inductor when inductors aredesigned. Table 1 shows the design condition of circuit
parameters and material parameters about circuit specificationand the inductor core. The Ferrite is selected for the material of
the inductor core. In addition, the shape of the inductor core is
EE type. The space factor is supposed from 0.4 to 0.6 at thedesign of the inductor. The boost inductor of the conventional
two stage type is designed by the ripple current value of theinductor. Therefore, it is designed by (11).
LVin VDC  Vin
DI L f sw VDC Sspace factor of wire
0.4~0.6

where Vin is the input voltage, VDC is the DC link voltage, fsw isthe switching frequency, DIL is the ripple current of theinductor at peak to peak. The interconnected inductor of theconventional two stage type is also designed by (11). Vin is thepeak voltage of the grid when the interconnected inductor isdesigned. In addition, fsw is two times compared with boost
inductor design due to the inverter is operated at PWM.
Fig. 10 shows design inductor values of the conventional
two stage type and the boost inverter. The boost inductor andthe interconnected inductor are 2.21 mH and 6.1 mH at the
input voltage is 48 V. On the other hand, the input inductor of
the boost inverter is 40.1 mH. The input inductor value of theboost inverter is reduced by 98.2% compared with the boost
inductor of the conventional two stage type. In addition, theeffect for minimization is large due to the conventional twostage type have the interconnected inductor without the boost
inductor. As a result, the boost inverter can reduce the circuit
volume compared with the conventional two stage type.
V.

lg

SIMULATION RESULTS

A. Operation of bosst inverter
Table 2 shows the simulation conditions, whereas Fig. 11
Table 2. Simulation conditions of boost inverter.

Material : Ferrite
Specification

Fig. 9. Designing inductor structure of each circuit toporogy.
Table 1 Design condition of inductors

Material condition ofInductor

Circuit condition
Design condition of
conventional circuit

Inductor value [mH]100

Shape type
Material
Initial permeability mi
Flux density Bs
Flux density Hc
Resistivity r
Output power Pout
Output voltage Vout
Switching frequency fsw
DClink voltage VDC
Ripple ratio of current DIL

Conventional circuit

EE coreN87
2200 ± 25%
490 mT
21 A/m
10 Wm
300 W
200 V
10 kHz
350 V
30 %

0

48 V

Switching frequency fsw

10 kHzBoost inductor

2.2 mHInterconnected inductor

6.1 mH

Boost inductor

40 mH

Output voltage Vout

100
200
Input voltage Vin [V]Output current Iout

Output capacitor voltage VCa,VCb

0.01

0.001

50 Hz

Inductor current IL

Input inductor(Prop.)

0.1

V

Output frequency fout

Input voltage Vin

Interconnected inductor(Conv.)

1

300 W

Input voltage Vin

Boost inverter

Boost inductor(Conv.)

10

Output power Pout
Output voltage Vout

300

Fig. 10. Comparison of designed indcutor values between conventional
two-satage type and boost inverter when the input voltage is changed.

Fig. 11. Operation each waveform of boost inverter circuit in simulation.

Fig. 12 shows the magnified waveform of the input
inductor current. It is confirmed the input inductor current is
operated in each period for the voltage of the output capacitor
VCa and VCb. As a result, the input current is realized DCM
operation in order to share the input inductor current for each
operation mode. In this simulation, the input inductor current
have the zero current period between the voltage control of VCa
and VCb.
Fig. 13 shows the neutral point voltage waveform of thegrid side at the operating boost inverter in simulation. The
neutral point voltage is kept the constant value by the averagevalue of the output capacitor voltage. Additionally, the voltage
ripple of the neutral point voltage is under the 10 V. As a result,the voltage changing of the grid side is suppressed due to thevoltage ripple of the neutral point voltage is low.
B. Loss analysis
Fig. 14 shows simulation results of the loss analysis. It is
compared with two DC/AC converters, the conventional twostage type and the boost inverter. In this case, the loss analysis
results include only the switching devices and the inductor.
The inductor loss is obtained from GeckoMAGNETICS of thesimulation software. From the results, the loss of the boost
inverter is reduced compared with the conventional two-stage

type. The switching device losses of the conventional type are
4.7 W. The loss of the boost inverter is 7.6 W. The switching
device losses of the boost inverter are 38.2% larger thanconventional two stage type. It is caused by the large effective
value of the input inductor current of the boost inverter
compared with the conventional two stage type. On the other
hand, inductor losses are 13.7 W of the conventional two stage
type due to the inductor is connected to boost converter sideand inverter side. In the boost inverter, the inductor loss is 6.6
W caused by connecting the small inductor. Therefore, theinductor loss of the boost inverter is 51.8% smaller than theconventional two stage type. As a results, the converter loss ofthe boost inverter can be reduced compared with theconventional two stage type due to the inductor current isshared in boost inverter.
Fig. 15 shows the RMS inductor current for each inductors.
These are the boost indcutor and the interconnected inductor of
the conventinal two stage type conpared with the input inductorof the boost inverter. The boost inductor current is 6.5 A at 48
V of the input voltage. Additionaly, the interconnectedinductor current is 1.5 A. On the other hand, the input inductor
DeviceConv. : GS66504B
Prop. : GS66516T

Efficiency
94.0%

20
16

Devices Loss [W]shows the simulation results of the boost inverter operation. It
is confirmed that the boost inverter can generate the sinusoidal
waveform of the output voltage when the input power supply is
DC source. In addition, the voltage of the output capacitor VCa
and VCb is controlled over the input voltage at each operation
mode periods. In consequence, the boost inverter is achieved
the boost mode and the step-down mode.

Inductorloss

12
Switching
loss

8
4

d3
d1

Conduction
loss

0

d4

d2

Efficiency
95.2%

Conventional
two-stage type

Proposedboost inverter

Fig. 14. Loss analysis of conventional circuit and proposed circuitin simulation.

Fig. 12. Magnifying operation waveform of input inductor current.

50

Output current Iout

Neutral point voltage Vn

Inductor current IL [A]

Output voltage Vout

40
30

Boost inductor(Conv.)

20

Interconnected inductor(Conv.)

100

Fig. 13. Measurement waveform of neutral point voltage
of proposed circuit.

Input inductor(Prop.)

Output power Pout : 300 W
Output voltage Vout : 200 V
Switching frequency fsw : 10 kHz
DC link voltage : 350V

0

200
100
Input voltage Vin [V]Fig. 15. Comparison of RMS of each inductro current.

300

current is 20 A. It is 67.5% large compared with the boost
inductor current. Therefore, it is confirmed the effective current
in the boost inverter is large compared with the convemtional
two stage type. In consecuence, the loss of the switching
device is inceresed in the boost inverter. However, the inductor
loss of the boost inverter is reduced due to the inductor voalueof the boost inverter is smaller than each indcutor of theconventional two stage type.
C. Inductor energy of boost inverter
Fig. 16 shows the inductor energy of each inductor of theconventional two stage type and the boost inverter. Energies ofthe boost inductor and the interconnected inductor are 43.2 mJ
and 6.9mJ. On the other hand, the input inductor energy is 8 mJ.
The input inductor energy is reduced by 84% compared withthe total inductor energy of the conventional two stage type.
Therefore, the inductor volume of the boost inverter is reduced
compared with the conventional two stage type. As a results,
The volume of the DC/AC converter can be reduced using theboost inverter.
VI.

EXPERIMENTAL RESULTS

Fig. 17 shows the experiment diagram of the boost inverter.
The experimental condition is connected only resistor load inorder to confirm the prototype system of the boost inverter. In

this case, the input inductor of the boost inverter is designed by(10). The input inductor value have 10% margin comparedwith the minimum design value in order to prevent the criticalmode when the condition is the rated power.
Fig. 18 and 19 shows each operation waveform of the boost
inverter when the output power is 100 W. It is confirmed thesinusoidal waveform of the output voltage at 100 V when the
input power supply is DC source. In addition, the output
current THD is measured 1.4%. The boost inverter is realized
the low current THD of the output current. On the other hand,each output capacitor voltage is kept the constant value of theaverage voltage due to voltage control in order to become overthe input voltage. Moreover, the output capacitor voltage is
Input voltage Vin [100 V/div]
Inductor current IL [20 A/div]000
Output voltage Vout [250 V/div]0
Output current Iout [2 A/div]

60

Total energy
50.0 mJ

Inductor energy [mJ]

50Interconnectedinductor

THD 1.4%

Time [10 ms/div]

Fig. 18. Experimental results of input and output waveforms.
Capacitor voltage VCa [100 V/div] Capacitor voltage VCb [100 V/div]

40

Reduced by
84%

30
20

Boost
inductor

10

Input
inductor0

Conventional
two-stage type

Inductor current IL [20 A/div]0

Total energy
8.0 mJ

Proposedboost inverter00
Output current Iout [1 A/div]

Fig. 16. Comparison of inductor energy between comventional two-stage
type and boost inverter.

Time [10 ms/div]

Fig. 19. Experimental waveform of each output capacitor voltage.

Device:SCT2080KE(Rohm)
Inductor current IL [10 A/div]

S3

L:30 mH

Rload
100 W

S4

0

S1

Vin:100 V

S2

S

Ca
fsw:20 kHz

Cb

10.5 mF

Fig. 17. Experimental circuit diagram using proposed circuit.

Time [20 ms/div]

Fig. 20. Magnifying DCM waveform of input inductor current.

oscillated at the output frequency. As a results, it is confirmed
the control of the boost inverter is operated.
Fig. 20 shows the magnified waveform of the inductor
current. The input inductor current is operated in DCM due tozero current period. Therefore, there are zero current period d0
between the voltage control of the output capacitor voltage VCa
and VCb. It is confirmed the input inductor current similar to
Fig. 6. In consequence, the input inductor current is shared
between each output capacitor voltage control in the boost
inverter. In addition, the inverter operation is realized by the
boost inverter confirming from experimental results.
VII. CONCRUSION
In this paper, the boost inverter using only one input inductor
is proposed applied for the micro-inverter in order tominimization of the volume. In addition, It is considered thedesign method of the input inductor, the simulation of thecircuit operation and the comparing the losses of converter
between the conventional two stage type and the proposed
boost inverter. As a results, it is confirmed the inductor loss ofthe proposed boost inverter is reduced by 51.8% comparedwith the conventional topology in the theoretical consideration.
Therefore, the operation of the boost inverter is confirmed inexperimental results. In consequence, the proposed boost
inverter can be converted DC to AC using only a input inductor.
In the future, the operation and the efficiency of the boost
inverter will confirm when the grid is connected at 300 W.
REFERENCES
[1]

[2]

[3]

[4]

S. Saridakis, E. Koutroulis, F. Blaabjerg : “Optimization of SiC-BasedH5 and Conergy-NPC Transformerless PV Inverters”, IEEE Journal of
Emerging and Selected Topics in Power Electronics, Vol. 3, No. 2, pp.
555-567 (2015)
Y. Zhou, Hongbo Li, Hui Li : “A Single-Phase PV Quasi-Z-Source
Inverter With Reduced Capacitance Using Modified Modulation andDouble-Frequency Ripple Suppression Control”, IEEE Transactions on
Power Electronics, Vol. 31, No. 3, pp. 2166-2173 (2016)
R. Chattopadhyay, S. Bhattacharya, N. C. Foureaux, I. A. Pires, H. dePaula, L. Moraes, P. C. Cortizio, S. M. Silva, B. C. Fil-ho, Jose A. de S.
Brito : “Low-Voltage PV Power Integration into Medium Voltage Grid
Using High-Voltage SiC Devices”, IEEJ Journal of Industry
Applications, vol.4, no.6, pp.767-775, (2015).
S. Saridakis, E. Koutroulis, F. Blaabjerg : “Optimization of SiC-BasedH5 and Con-ergy-NPC Transformerless PV Inverters”, IEEE Journal of
Emerging and Selected Top-ics in Power Electronics, vol. 3, no. 2, pp.
555-567, (2015).

[5]

S. Yamaguchi, T. Shimizu : “Single-phase Power Conditioner with a
Buck-boost-type Power Decoupling Circuit”, IEEJ J. Industry
Applications, vol.5, no.3, pp.191-198, (2016).
[6] C. Liao, W. Lin, Y. Chen, C. Chou : “A PV Micro-inverter With PV
Current Decoupling Strategy”, IEEE Trans. on Power Electronics, Vol.
32, No. 8, pp. 6544-6557 (2017).
[7] M. A. Rezaei, K. Lee, A. Q. Huang : “A High-Efficiency Flyback
Micro-inverter With a New Adaptive Snubber for Photovoltaic
Applications”, IEEE Trans. on Power Electronics, Vol. 31, No. 1, pp.
318-327 (2016)[8] A. C. Nanakos , G. C. Christidis , E. C. Tatakis : "Weighted Efficiency
Optimization of Flyback Microinverter Under Improved Boundary
Conduction Mode (i-BCM)", IEEE Transactions on Power Electronics,
Vol. 30, No. 10, pp. 5548-5564 (2015)[9] W. J. Cha, Y. W. Cho , J. M. Kwon , B. H. Kwon : "Highly Efficient
Microinverter With Soft-Switching Step-Up Converter and SingleSwitch-Modulation Inverter", IEEE Transactions on Industrial
Electronics, Vol. 62, No. 6, pp. 3516-3523 (2015)[10] D. Meneses , O. García , P. Alou, J. A. Oliver, J. A. Cobos : "GridConnected Forward Microinverter With Primary-Parallel SecondarySeries Transformer", IEEE Transactions on Power Electronics, Vol. 30,
No. 9, pp. 4819-4830 (2015)[11] M. Jang, V. G. Agelidis : “A Minimum Power-Processing-Stage FuelCell Energy System Based on a Boost-Inverter with a Bidirectional
Backup Battery Storage”, IEEE Trans. on Power Electronics, Vol. 26,
No. 5, pp. 1568-1577 (2011).
[12] R. O. Caceres, I. Barbi : “A boost DC-AC converter: analysis, design,and experimen-tation”, IEEE Transactions on Power Electronics, vol.
14, no. 1, pp. 134-141, (1999).
[13] D. B. W. Abeywardana, B. Hredzak, V. G. Agelidis : “An Input Current
Feedback Method to Mitigate the DC-Side Low-Frequency Ripple
Current in a Single-Phase Boost Inverter”, IEEE Transactions on PowerElectronics, vol. 31, no. 6, pp. 4594-4603, (2016).
[14] W. Zhao, D. D. C. Lu, V. G. Agelidis : “Current Control of GridConnected Boost Inverter With Zero Steady-State Error”, IEEE
Transactions on Power Electronics, vol. 26, no. 10, pp. 2825-2834, .
(2011).
[15] Y. Tang, Y. Bai, J. Kan, F. Xu : “Improved Dual Boost Inverter With
Half Cycle Modulation”, IEEE Transactions on Power Electronics, vol.
32, no. 10, pp. 7543-7552, (2017).
[16] T. Morita et al. : “650 V 3.1 mWcm2 GaN-based monolithic bidirectionalswitch using normally-off gate injection transistor”, 2007 IEEE
International Electron Devices Meeting, Washington, DC, 2007, pp.
865-868.
[17] H. N. Le, K. Orikawa, J. I. Itoh : “Circuit-Parameter-Independent
Nonlinearity Compensation for Boost Converter Operated in
Discontinuous Current Mode”, IEEE Transactions on Industrial
Electronics, vol. 64, no. 2, pp. 1157-1166, (2017).