200-Gb/s waveform analysis ofultrafast all-optical semiconductor gates
towards low-power consumption operation

Jun Sakaguchi

The Twenty-First Century Center of Excellence Program“Innovation in coherent optical science”
Graduate School of Electronic Engineering
University of Electro-Communications
Doctor of Philosophy
June 2008

200-Gb/s waveform analysis ofultrafast all-optical semiconductor gates
towards low-power consumption operation

Approved by supervisory committee:
Chairperson: Assoc. Prof. Yoshiyasu Ueno
Member: Prof. Takeshi Kamiya
Member: Prof. Kenichi Ueda
Member: Assoc. Prof. Tsuyoshi Okuno
Member: Assoc. Prof. Naoto Kishi
Member: Prof. Kouichi Yamaguchi
Approved by external reviewers:
Prof. Harm J. S. Dorren
Prof. Jesper Mørk
Prof. Jürg Leuthold.

Copyright
byJun Sakaguchi
2008

Abstract
This thesis is aimed at a fundamental understanding of all-optical signal processing gates based on semiconductor optical ampliﬁers (SOAs), for their low-power
consumption operation in ultrafast frequency region. In particular, modeling of afrequency-dependent electric power consumption in the SOA and suppression of several kinds of output waveform degradations in the ultrafast wavelength conversion
are discussed in detail.
After an introduction in Chapter 1, we present a power consumption model of an
SOA gate in Chapter 2, to clarify the power consumption dependence on the operation frequency B and other fundamental parameters. The carrier density dynamics
in the SOA was modeled using a rate equation including three diﬀerent carrierconversion eﬃciencies of the SOA, which we introduced as the dominant factors.
The validity of this rate equation model was tested with measured characteristics
of custom-designed SOA samples with diﬀerent structures. Using the model, we
predicted that the power consumption increase with B 2 in the high frequency limit.
This B 2 -dependence was experimentally demonstrated for two diﬀerent SOA samples in the frequency range of 20 ∼ 100 GHz. We expect that this initial model willcontribute to the designing of SOA power consumption or to the future outlook of
SOA gates.
In Chapter 3, we discuss the generation of small sub-pulses from a delayed interference signal-wavelength converter (DISC). This sub-pulse generation is regarded
as one potential issue which prevents the practical operation of the DISC. We experimentally revealed the sub-pulse generation in a relatively low frequency region(10 ∼ 25 GHz) through cross-correlation measurements, and veriﬁed our sub-pulse
generation model with high accuracy (1 ∼ 2 dB). We demonstrated the increaseof sub-pulse intensity up to −10 dB with the carrier recovery rate, which will leadto the trade-oﬀ between the sub-pulse intensity and the pattern-induced intensity
ﬂuctuation. The sub-pulse model veriﬁed in this work will be valuable for future
solution of this issue.
In Chapter 4, we report on our wavelength conversion experiment for 200 Gb/s,
4992-bit data signal using a DISC gate. For this purpose we developed an original

low jitter, low repetition rate (40 MHz), ultrafast (600 ∼ 700 fs) and synchronized sampling pulse generator for the cross-correlator input and succeeded in precise waveform-monitoring of the ultrafast long-pattern signal. The pattern-induced
intensity ﬂuctuation (PIF) of the output waveform was systematically measured,and the impact of the holding-beam injection and bandpass ﬁlter detuning on its
suppression were compared. We systematically exhibited that the ﬁlter detuning
eﬃciently suppressed PIF even for 200 Gb/s, long-pattern data signal and it saved
the required holding-beam power by ∼ 3 dB. We also discovered a new method
using the nonlinear polarization rotation in the SOA, and obtained a guideline foruse of this method thorough systematic measurements. These results will provide
indispensable backgrounds for the practical, ultrafast, low-power operation of the
DISC.
In conclusion, we developed the understanding of the SOA gate (DISC) operated
in the frequency range of 10 ∼ 200 Gb/s, and considerably facilitated its low-power,
small signal-degradation operation and the system design.

5

CONTENTS

Contents
List of acronyms used in this thesis

8

1 Introduction

10

2 Frequency-dependent electric power consumption model including
quantum-conversion eﬃciencies in ultrafast SOA gates15
2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.2 Our model of carrier conversion eﬃciency . . . . . . . . . . . . . . . . 18
2.2.1 Deﬁnition of carrier conversion eﬃciency . . . . . . . . . . . . 18
2.2.2 Characterization of the conversion eﬃciency . . . . . . . . . . 19
2.3 Conversion eﬃciency measurement and results . . . . . . . . . . . . . 21
2.4 Estimation of SOA electrical power consumption of all-optical gating 32
2.4.1 Rate equation model including conversion eﬃciencies . . . . . 32
2.4.2 Electrical power consumption . . . . . . . . . . . . . . . . . . 35
2.4.3 High-frequency limit . . . . . . . . . . . . . . . . . . . . . . . 38
2.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
3 Observation of sub-pulse generation from the DISC-type wavelength converter
42
3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
3.2 Sub-pulse generation model . . . . . . . . . . . . . . . . . . . . . . . 43
3.2.1 DISC-gate Operation model based on the SOA Carrier Dynamics 43
3.2.2 Sub-pulse generation model . . . . . . . . . . . . . . . . . . . 46
3.3 Experimental Setup for Sub-pulse Measurement . . . . . . . . . . . . 46
3.3.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
3.3.2 Phase bias of polarization-based MZI . . . . . . . . . . . . . . 52
3.4 Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

6

CONTENTS

3.4.1

Sub-pulse measurement and calculation with systematic MZIbias shifts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
3.4.2 Trade-oﬀ regarding sub-pulse generation and pattern-induced
intensity ﬂuctuation . . . . . . . . . . . . . . . . . . . . . . . 59
3.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
4 200-Gb/s wavelength conversion using DISC with long-pattern waveform monitoring64
4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64
4.2 Experimental setup for the 200-Gb/s wavelength conversion . . . . . 65
4.2.1 Optical time-division multiplexing of mode-locked pulses up
to 200 GHz for the gate input . . . . . . . . . . . . . . . . . . 69
4.2.2 Generation of low-jitter sampling pulses at 40 MHz . . . . . . 73
4.2.3 Waveform monitoring of long-pattern 200-Gb/s optical signal . 78
4.3 Wavelength conversion with DISC at 200 Gb/s . . . . . . . . . . . . . 81
4.3.1 Impact of the holding-beam eﬀect . . . . . . . . . . . . . . . . 83
4.3.2 Impact of the ﬁlter detuning . . . . . . . . . . . . . . . . . . . 86
4.3.3 Impact of the nonlinear polarization rotation . . . . . . . . . . 90
4.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
5 Conclusions and outlook
100
5.1 Conclusions of this thesis . . . . . . . . . . . . . . . . . . . . . . . . . 100
5.2 Future outlook . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102
A Entire waveform of the 200-Gb/s input signal

105

Acknowledgments

110

References

112

List of publications

123

7

List of acronyms

List of acronyms used in this thesis
AL
Assist Light
ASE
Ampliﬁed Spontaneous Emission
BPF
Bandpass Filtercw
continuous wave
DCF
Dispersion Compensation Fiber
DDF
Dispersion Decreasing FiberDFB
Distributed Feedback
DGD
Diﬀerential Group Delay
DISC
Delayed-Interference Signal-wavelength Converter
DSF
Dispersion Shifted Fiber
EA
Electro-Absorption
EDFA
Erbium-Doped Fiber Ampliﬁer
EO
Electro-optic
FP
Fabry-Perot
GVD
Group Velocity Dispersion
HNLF
Highly Nonlinear Fiber
ISBTIntersubband transitionLD
Laser Diode
LN
Lithium Niobate
MLFL
Mode-Locked Fiber Laser
MZI
Mach-Zehnder Interferometer
MQW
Multiple Quantum Well
NPR
Nonlinear Polarization Rotation
OE
Opto-electronic
OSA
Optical Spectrum Analyzer
OTDM
Optical Time-Division Multiplexing
PBS
Poralization Beam Splitter
PC
Polarization Controller
PD
PhotoDiode
PD-SMZ Polarization-Discriminating Symmetric Mach Zehnder
PIF
Pattern-induced Intensity Fluctuation
PMPolarization Maintaining

8

List of acronyms

PPG
Programmable Pattern Generator
PRBS
Pseudorandom Bit Sequence
RFRadio Frequency
RZ
Return-to-Zero
SCH
Separate Conﬁnement Heterostructure
SFG
Sum Frequency Generation
SHG
Second Harmonic Generation
SOA
Semiconductor Optical Ampliﬁer
SMF
Single-Mode Fiber
SMZ
Symmetric Mach Zehnder (structure)
SPM
Self Phase Modulation
SSG
Small Signal Gain
TE
Transeverse Electric
TM
Transeverse Magnetic
TOAD Terahertz Optical Asymmetric Demultiplexer
UNI
Ultrafast Nonlinear Interferometer
VDL
Varriable Delay Line
VOA
Varriable Optical Attenuator
XGM
Cross Gain Modulation
XPM
Cross Phase Modulation
WDM
Wavelength-Division Multiplexing

9CHAPTER 1. INTRODUCTION

Chapter 1
Introduction

The all-optical signal processing has been drawing many researchers’ interest for along time, as a method to overcome the frequency limitation of conventional signal processors. While the operation frequencies of conventional, electric processors
are limited by their gate capacitances and load resistances, the all-optical signal
processing potentially works at as high speed as that of the optical nonlinear phenomena involved[1]. As the development of highly nonlinear devices reduced theoptical signal power required to cause such nonlinear eﬀects, the all-optical signal processing has become recognized as an energy-eﬀective solution for high-speed
data-communication systems.
The optical time-division multiplexing (OTDM) [2–4] network is an example ofsuch solution, and supposed to be a cost-eﬀective alternative for the wavelengthdivision multiplexing (WDM) technology. The latter, schematically shown in Fig.
1.1(a), has made considerable success in large-capacity data transmission, indeed.
Many data signals with relatively low bit-rate were multiplexed in the wavelength
space (together with polarization-multiplexing and diﬀerential quadrature phaseshift keying), and the aggregated transmission rate reached 25.6 Tb/s[5]. It is
not cost-eﬀective, nevertheless, to build a complicated network including functions
as routing or signal regeneration, based on the WDM technology. Thus we haveaimed to introduce OTDM to the WDM-based network, as schematically shownin Fig. 1.1(b). The function block in the ﬁgure is drastically simpliﬁed owing
to frequency-transparent all-optical signal processors, compared to that in the lowfrequency WDM network. In such OTDM-WDM network, many functions as signalwavelength conversion, 3R regeneration (re-ampliﬁcation, re-shaping, re-timing),
OTDM demultiplexing, header recognition, simple logic-operation, and buﬀering

10CHAPTER 1. INTRODUCTION(a)function block
Conventional
xM
Signal processors
WDM
MUX
M channels
x N b/s(N: 10~40 G)

(b)
WDM
MUX

WDM
MUX

WDM
DEMUX

All-opticalx M’
Signal processors
WDM
WDM
MUX
DEMUX

WDM
DEMUX

WDM
DEMUX

M’ channels
x N’ b/s(N’ : ~160 G)

Figure 1.1: Schematic drawings of signal processing (a): that in WDM networks,
using conventional signal processors. (b): that in OTDM-WDM networks, using
all-optical signal processors.
are desired to be realized by all-optical signal processors. Thus many kinds of alloptical gates for these functions have been intensively studied.
Among several kinds of all-optical signal processing, those using the cross-phasemodulation (XPM) in the carrier-injected semiconductor optical ampliﬁer (SOA)[6]have been the most practical candidate[7–9]. After the ﬁrst demonstration of the
Terahertz Optical Asymmetric Demultiplexer (TOAD) in 1993[10], many kinds of
XPM-based SOA gates such as Symmetric Mach-Zehnder (SMZ, Fig. 1.2(a))[11, 12],
Polarization-Discriminating Symmetric Mach-Zehnder (PD-SMZ, Fig. 1.2(b))[13,14], or Delayed-Interference Signal-wavelength Converter (DISC, Fig. 1.2(c))[15]
have been proposed and demonstrated. These interferometric SOA-gates are advantageous against optical gates based on Kerr-nonlinearity in an optical ﬁber interms of control pulse energy, environmental stability, polarization sensitivity, and11

CHAPTER 1. INTRODUCTION

integrability[16–19]. They are also advantageous against XGM-based SOA gates interms of chirp characteristics and signal extinction ratio. Research eﬀorts mostly
from commercial companies have been devoted to the high-frequency operation andpractical application of these gates, and all-optical demultiplexing at 336 Gb/s with
SMZ[20], 3R regeneration at 84 Gb/s with PD-SMZ[21, 22]∗) , and wavelength conversion at 160 Gb/s with DISC[23, 24] have been demonstrated. Even 320- to 640GHz operation of all-optical SOA gates using ultrafast chirp dynamics has recently
been demonstrated[25–27]. Among these gate functions, the wavelength conversion[28] has been one of the most developed and successful functions. SOA-based
wavelength converters have been already used as indispensable components in morepractical experiments[29, 30].
While many practical demonstrations have been successfully achieved, fundamental understanding of these SOA-gates has not been well advanced especiallyin the frequency range of over 100 Gb/s. Particularly, minimum requirements on
the operating condition of these gates have been unknown. It has been diﬃcult
to answer how much electric power is essential for the gating operation at certain
frequency. This question is practically important in realizing highly-integrated SOA
based optical circuits in the future or to reduce the power consumption of the wholesystem. Also it has been practically recognized that many other factors as well as
the electric power consumption aﬀect the quality of the output signal from SOA
gates. One common degradation of the signal is the pattern-induced intensity ﬂuctuation (PIF), which results from imperfect recovery of the carrier depletion inside
the SOA. Well-known solution for that is an injection of a strong cw light into the
SOA (holding-beam eﬀect, [31], or assist-light eﬀect [32–39]). Also detuning of the
bandpass ﬁlter (BPF) after the SOA [40–43], or similar and advanced approach [44],are known to be eﬀective in the suppression of PIF. These techniques contributed tothe demonstration of the past ultrafast gating for practically-short (27 − 1) length
data signal. As we found through the experiment, nonlinear polarization rotation(NPR) in the SOA can be also used for its suppression. Systematic evaluation andcomparison of these eﬀects in the frequency range of ∼ 200 Gb/s (and long data pattern) has not been reported, however, to the author’s knowledge. For other kinds ofwaveform degradation as sub-pulse generation, hardly anything has been studied in∗)

The assessment of the regeneration property of SOA delayed interference conﬁguration (SOADI) at 160 Gb/s was also reported[24]. The structure of the SOA-DI is identical to DISC, whereas
its operation condition principle was diﬀerent in two senses: The delay time ∆t in the MZI was
almost equal to the bit spacing (6.25 ps). 2.1-ps clock signal was used instead of the cw light,
which determined the output pulse width rather than the MZI.

12

CHAPTER 1. INTRODUCTION(a)control pulse

Semiconductor
OpticalAmplifier (SOA)
output signal
phase shifter

input signal

bandpass filter∆T-delayed control pulse(b)phase shifter

control pulse

output signal∆T-delay
Polarizer

input signal∆T-delay(c)phase shifter

input signal

output signal

cw light∆T-delay∆T(d)control pulse

output signal
input signal

Figure 1.2: Various structures of XPM-based SOA gates. (a): Symmetric MachZehnder (SMZ) (b): Polarization-Discriminating Symmetric Mach-Zehnder (PDSMZ), or Ultrafast Nonlinear Interferometer (UNI) (c): Delayed-Interference Signalwavelength Converter (DISC), and (d): Terahertz Optical Asymmetric Demultiplexer (TOAD).

13

CHAPTER 1. INTRODUCTION

detail. Characteristics of these degradation must be clariﬁed to design and actuallyoperate the all-optical signal-processing system based on the SOA gates.
In this thesis, we discuss these issues both theoretically and experimentally.
In Chapter 2, we explain our development of a practical model that scientiﬁcallydescribes the dc power consumption of an SOA-based all-optical gate as a function of
its working frequency up to 160 GHz. In Chapter 3, we discuss our ﬁrst observationof the sub-pulse generation out of the DISC gate, and its theoretical analysis. In
Chapter 4, we discuss our results on waveform measurements of the DISC gate at 200
Gb/s, measured PIF, and several methods to suppress it. These research activities
will contribute to the practical, low-power consumption, and ultrafast operation of
the DISC and other SOA gates, and the realization of the future OTDM-WDM
networks. In Chapter 5, we discuss the overall conclusion of this thesis and its
impact to the other activities.

14

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Chapter 2
Frequency-dependent electric power consumption
model including quantum-conversion eﬃciencies
in ultrafast SOA gates

2.1

Introduction

As mentioned in Chapter 1, all-optical signal-processors based on the XPM in SOAs
are promising for future OTDM networks, for their capability of ultrafast signal processing and low power consumption. The SOA gates used in the past demonstrationof ultra-high frequency demultiplexing, 3R regeneration, and wavelength conversion
typically consumed 0.4− to 1.0−W electrical power, while the latest electrical demultiplexer consumes 5.5 W in 100-GHz operation[45]. Even though these SOA
gates need additional power consumption for stabilizing the operating temperature(0.1 ∼ 1 W in the ideal cases (Fig. 2.1) and more depending on the system coolingeﬃciency), they should be much advantageous compared to their electric counterparts. As the capabilities of SOAs for ultrafast gating have become clear, the lower
limit of electric power consumption and its origin have become important design
issues. A practical engineering model that describes dc power consumption has,
however, not yet been reported to the best of the authors’ knowledge. Because sucha model has not been available, it has been very diﬃcult for system researchers
to design the total power consumption of the various large-scale 160-GHz OTDM
systems now under research.
The electrical power consumption of an SOA in an all-optical gate is dominatedby two factors: the amount of optical-pulse-induced modulation as the optical phaseshift in the case of XPM, and the recovery time of the depleted carrier density after
optical modulation. It has long been recognized that the so-called optical holding
15

Power consumption in TEC (W)

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES100

G =Area/thickness(mm)
18(used in this work)
4.5
8.0
32
53
64
111
571

10
1
0.1

y~x 3
0.01
0.01 0.1

1

10

100

TEC cooling power (W)Figure 2.1: Ideal power consumptions of several commercial thermoelectric coolers asfunctions of their cooling powers, calculated from speciﬁcation data of each product.

beam eﬀectively accelerates the carrier’s recovery time [31–39], while the holding
beam itself consumes a signiﬁcant part of the injected carriers via the stimulated
recombination process and consequently reduces the amount of optical phase shift
in the co-propagating optical signal component. It has also been qualitatively wellknown that to accelerate the recovery time while maintaining the amount of opticalphase shift in the 0.3- to-1.0 π range, the carrier injection rate (i.e., the injection
current density) must be increased, and this increases in the dc power consumption.
To discuss the power consumption, a detailed understanding of the quantum
conversion eﬃciency from the number of injected carriers to the number of opticalpulse-induced stimulated recombinations in the SOA is needed. It is reasonably
expected that an SOA with low conversion eﬃciency requires a large current injection for the gating operation. Furthermore, an increase in the injection current
density might slightly degrade the quantum conversion eﬃciency. This quantum
eﬃciency was not taken into account in most previous studies. Some researchers
discussed SOA gain within continuous wave (cw) regime and carrier recovery using
one conversion eﬃciency parameter[46, 47], but no one has studied how conversion
eﬃciency is inﬂuenced by the SOA properties. Furthermore, no study of the conversion eﬃciency in ultrafast optical gating has been reported, even though this hasturned out to be signiﬁcantly lower than that for cw-light ampliﬁcation through16CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

present research.
In this chapter, we explain our development of a practical engineering model thatscientiﬁcally describes the dc power consumption of an SOA-based all-optical gate
as a function of its working frequency up to 160 GHz. A model of the holding beam’s
contribution, which is a part of our power-consumption model, is veriﬁed througha series of measured results. In these veriﬁcation processes, measured parametervalues are put into most of the independent model parameters, such as several
types of quantum eﬃciencies. The observed dependence of the measured quantum
eﬃciencies on the SOA structural parameters are also discussed.

17

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

2.2

Our model of carrier conversion eﬃciency

We deﬁne three diﬀerent carrier conversion eﬃciencies, and use them to evaluate
diﬀerent carrier losses in the SOA. Figure 2.2 shows the loss model for injected
carriers.

2.2.1

Deﬁnition of carrier conversion eﬃciency

We start from the total carrier number N and carrier density n in the active region.
When n is below the population inversion threshold n0 , the SOA has a negative
gain and shows no band-ﬁlling nonlinearity. Thus, the injection current Iop should
be larger than the transparent current I0 (where n = n0 ) and the number of carriers
available for the gating should be reduced to the excess carrier number Nex , where
Nex = N − N0 . For this process, we deﬁne the ﬁrst conversion eﬃciency η1 as
η1 = Nex /N .
Next, we introduce the carrier number Ncw , which is the maximum number ofstored carriers available for ampliﬁcation of saturating cw light. This can also be
understood as the number of carriers available for optical gating at the low frequency
limit. Ncw is supposed to be smaller than Nex deﬁned above, because of the eﬀects ofASE, Auger recombination, carrier overﬂow, intervalence band absorption, and free
carrier absorption. The second conversion eﬃciency η2 is deﬁned as η2 = Ncw /Nex .
In this paper, we will not discuss the individual loss contributions, because of theexperimental diﬃculty of determining these.
Finally, we introduce the carrier number Npulse , which is the maximum numberof stored carriers available for ampliﬁcation of ultrafast optical pulses. Npulse issupposed to be smaller than Ncw , and the third eﬃciency η3 is deﬁned as η3 =Npulse /Ncw . Its potential causes are spectral hole burning and carrier heating[49–53], or increases in ASE recombination caused by the higher average-carrier density
compared with the cw-light ampliﬁcation case, and so on. Note that this diﬀerencein SOA reactions to cw light and ultrafast pulses only appears well above the small
signal regime, where practical gating operation is carried out. We consider Npulse tohave the largest inﬂuence on ultrafast gating performance. Therefore, we deﬁne thetotal carrier-conversion eﬃciency ηT for ultrafast gating as ηT = Npulse /N = η1 η2 η3 .
If the dominant factors of each eﬃciency can be revealed, we will be able to maximize
the total eﬃciency by improving each contribution separately.

18

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES(a) Injection current Iop

η1
η2(Current at transparency, I0 )(b) Current over transparent condition, Iop-I0
ASE emission, Auger recombination

η3 (c) carriers which contribute to cw light amplification(hole burning, increase in ASE, etc. )(d) carriers which contribute to pulse light amplification
Figure 2.2: Loss model of injected carriers. η1 to η3 stand for conversion eﬃciencies.

2.2.2

Characterization of the conversion eﬃciency

We suppose that the carrier numbers and conversion eﬃciencies deﬁned above canbe characterized from fundamental SOA parameters as follows. When the carrierloss rate is 1/τC (τC represents the carrier lifetime) and the total carrier number
is N , the carrier loss N/τC should balance the injection ﬂow Iop /q (q: elemental
charge). Thus, the total carrier number N can be estimated as
N=

Iop
τC
q

(2.1)

η1 and the excess carrier number are supposed to be given by the transparency
current I0 as
η1 =

Nex
Iop − I0
=
N
Iop

(2.2)

Here, we should mention that N and Nex obtained from these relations contain certain underestimations when the dependence of the carrier loss rate on the injection
current is large. We will discuss it in §2.3.
Ncw is supposed to be obtained from the gain-saturation measurement of the
SOA for cw light. As the intensity of the input light increases, the SOA chip gain
G decreases from the small signal gain (SSG) G0 . For an SOA with saturationsat
and G0 À 1, G is suppressed by 3 dB when the output power reaches
power Pcw
sat
P3dB = Pcw · ln 2[55]. Taking the conventional discussion [54, 55] regarding the gain

19

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Table 2.1: List of SOA samples and their structure
Active region Active region Active region Conﬁnement
Sample Typelength,width,
thickness,factor,Leﬀ (µm)
wA (µm)dA (µm)
Γ
A#1
Bulk
1000
2
0.1
0.1 ∼ 0.2
A#2
Bulk
700
2
0.1
0.1 ∼ 0.2
A#3
Bulk
300
2
0.1
0.1 ∼ 0.2
A#4
Bulk
700
2
0.1
0.1 ∼ 0.2
∗1
∗2
B#1
MQW
1100
1.25
0.038
0.2
B#2
MQW
700∗1
1.25
0.038∗2
0.2
∗1
∗2
B#3
MQW
500
1.25
0.038
0.2
∗1
∗2
B#4
MQW
1100
1.25
0.038
0.2
C#1
Bulk
700
1.2
0.2
0.3 ∼ 0.4
D#1
Bulk
1500
0.6
0.35
0.5
(*1) Eﬀective length, considering 50 % contribution from taper regions(*2) Sum of well thickness
sat
saturation into account, we relate Ncw to Pcw, Gcw
0 (G0 for the cw light) and τC asη2 =P sat τC
Ncw
, Ncw = cw
ln Gcw
0 .
Nex
hν

(2.3)

Similarly, Npulse is supposed to be obtained from the gain-saturation measurement of the SOA for ultrafast pulses. If the injected pulse train has a much longer
interval than τC , the net gain of the pulse will be as follows[55]:Gpulse (Eout ) =

Eout
1
(
)
sat
sat
Epulse ln{exp Eout /Epulse + G0 − 1} − ln (G0 )

(2.4)

sat
The pulse saturation energy Epulse
can be obtained from the 2.35-dB suppression
pulse
point, and we relate it and G0(G0 for the ultrafast pulse) to Npulse as follows:

η3 =

sat
EpulseNpulse, Npulse =ln Gpulse
.
0
Ncw
hν

(2.5)

The intrinsic diﬀerence between Ncw and Npulse seems to be overlooked in the conventional gain-saturation theory. In the new rate equation model (see §2.4.1), bothcw and pulse saturation are modeled as independent characteristics.

20

ASE power from chip (dBm/nm)

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

0
-20
-40
A#1
A#2
A#3
A#4

-60
-80

B#1
B#2
B#3

-100

C#1, 150 mA
D#1

1.2 1.3 1.4 1.5 1.6 1.2 1.3 1.4 1.5 1.6 1.2 1.3 1.4 1.5 1.6 1.7

Wavelength ( µm)

Figure 2.3: Measured ASE spectra of SOA samples at Iop = 200 mA.

2.3

Conversion eﬃciency measurement and results

To evaluate the carrier conversion eﬃciency of actual SOAs, we prepared several
custom-designed SOA chips and commercial modules from diﬀerent manufacturers(A to D) and measured their fundamental parameters. Attributes of the SOA samples and examples of ASE spectra are shown in Table 2.1 and Fig. 2.3, respectively.
Most notable classiﬁcation of the SOAs is whether they have bulk structures or multiple quantum well (MQW) structures. While MQW structures have shown many
characteristics advantageous for the LD application (low propagation loss, small
threshold current, low α parameter, high modulation frequency), the superiority of
MQW SOAs for the xpm-based gating application has not been demonstrated, tothe author’s knowledge. Strained, polarization-insensitive MQW SOAs have often
been believed to be unsuitable, on the contrary, because their active layer thicknesses
were limited below the critical thicknesses and large conﬁnement factors cannot be
achieved as a consequence. Therefore the direct comparison of bulk and MQW
structures should be meaningful. For series A and B, we prepared chips with diﬀerent (eﬀective) active-region lengths Leﬀ . (The structural parameters were deﬁned as
schematically shown in Fig. 2.4.) The B#4 sample is the SOA used throughout the
200-Gb/s experiment in §4 and will provide good comparison with B#1 sample. For
each custom-designed chip, except for the shortest one (A#3), more than two wires
per electrode were bonded as shown in Fig. 2.5(a) to enable injection currents over
500 mA. We manufactured SOA-chip mounts with temperature controllers in viewof the dimensions of supplied chip submounts (one example shown in Figs. 2.5(b)21

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

yy
y
yyy
yyy
yyy

(a)

(b)

p-InP

Active region n-InP(InGaAsP,
λg~1.55 µm)

SCH (InGaAsP)

dAbarrierTM

n-InP substrate

(c)

well
Guiding layerTE(d)AR-coating
Waveguide
Taperwindow

wA

wA
L

L3
L2

L1

Leff = 12 L1+L2+ 12 L3
Figure 2.4: Drawing of typical SOA-structures and deﬁnitions of dimensional parameters. (a): Cross-sectional view, Bulk active region. (b): MQW active region.
(c): Top view, without taper and window regions. (d): Top view, with taper andwindow regions.

22

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

to (d)). Temperatures were stabilized at close to 250 C during the measurements. It
was unfortunate that the cooling eﬃciencies of these mounts were not quite good,
and the heat from the SOA chips sometimes caused thermal expansion to the whole
mount. Therefore experiments with larger injection current than 400 ∼ 500 mA
were diﬃcult due to rapid decreases in the chip-ﬁber coupling eﬃciencies.
The typical setup for SOA-chip characterization is shown in Fig. 2.6. We coupled
lensed ﬁbers to the input and output facets of the chip using precision stages (asshown in Figs. 2.7(a) and (b)), and obtained 2- to 5-dB coupling losses for each facet.
To accurately control the polarization angle of the light in the later chapters, weused a polarization-maintaining dispersion-shifted (PM-DS) lensed ﬁber fabricatedby Namiki Precision Jewel co. ltd. (Fig. 2.7(c)). The ASE intensities from both
facets were continuously monitored by power meters, and through careful adjustmentof the couplings ﬂuctuations were kept below 0.2 dB.
First we measured the cw-gain using a DFB-LD (λ2 = 1548.5 nm) and an optical
spectrum analyzer, from careful comparison between the spectral intensities of the
input and output light. Figure 2.8 plots the measured SSG of each sample against
Iop . The transparent currents I0 were obtained from these results. Figure 2.9(a)satshows typical gain-saturation proﬁles. We obtained Pcw
as a function of Iop fromsat
the 3-dB suppression points. As can be seen in Fig. 2.10(a), Pcw
increased almost
linearly with Iop without convergence. The SOA gains for ultrafast pulses were
measured for each sample. To obtain ultrafast optical pulses with lower frequency
than 1/τC , we used a mode-locked ﬁber laser (MLFL: Pritel Inc., UOC-3) with an
external LiNbO3 modulator. Pulses with a 2-ps width, λ1 = 1555 nm, and 10.5GHz frequency from the MLFL were modulated down to a 0.65-GHz pulse train.
We monitored the auto-correlation trace of the pulse train so that we could keep
its extinction ratio above 20 dB. Typical gain-saturation proﬁles are shown in Fig.
2.9(b). We see that for each Iop , the SSGs for the cw input are almost the same
as for ultrafast pulses. We ﬁtted the measured proﬁles to Eq. (2.4), and they
showed good agreement up to Eout ∼ 3000 fJ. The diﬀerence at Eout > 3000 fJ wasprobably caused by the slight residual of the 10.5-GHz pulses. From the ﬁt results,sat
we obtained Epulseof each SOA as a function of Iop (Fig. 2.10(b)). In contrast to
sat, they converged as Iop increased.
the case of Pcw
The carrier recovery rates 1/τC of each SOA at several injection currents were determined from XGM measurements using a cross-correlator (Femtochrome ResearchInc., FR-103XR) and mode-locked pulses modulated down to 1.3 GHz. Externalvariable delays and an elaborate calibration process were used to expand the scan
23

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

(a)

A#2

A#3

Leff = 1000 µm

Leff = 700 µm

Leff = 300 µm

B#1

B#2

B#3

Leff = 1100 µm

Leff = 700 µm

Leff = 500 µm(2 array)(2 array)

A#1

(b)
IOP

(c)

SOA chip submount
SOA chip

Thermister

pertiercoolerOxygen-free copper

IOP

submount
Lensed fiber

Teflon@Aluminum(d)Figure 2.5: Photographs of the SOA samples and drawings of the SOA mounts.
(a): Microscopic images of the custom-designed SOA samples. (b): The mount for
A-series SOA chips, side view. (c): Top view. (d) Photograph.
24

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATESCW light
λ2=1548 nm

VOASpectrum
analyzerPC

2 ps, λ1=1555 nm,
0.65~2.6 GHz

PC

VOA

lensed fiber(non PM)(PM)(Gain measurement)

SOA chipSpectrum
analyzer

PBS

Pol BPF
λ1, 5nmPowermeter (coupling monitor)

Pol
PC
2 ps,
10.5 GHz
MLFL

Power
meter

Power
meter

DFB LD

LN(coupling monitor)PC
1/4~1/16

clock
Pattern
Generator

BPF
λ1, 5nm(XGM)
Pol PCPC

Auto (cross)correlator

delay
BPF
λ2, 5nmBPF
λ2, 3nm10.5 GHz

Figure 2.6: Experimental setup for SOA chip characterization. VOA: variable optical attenuator, Pol: polarizer, PC: polarization controller, PBS: polarizing beam
splitter, BPF: band-pass ﬁlter, MLFL: mode-locked ﬁber laser
range to 600 ps. The intensity of the probe cw light was typically set to −25 dBm tosuppress the holding beam eﬀect. Pulse energies into each SOA chip were adjusted
within 10 ∼ 500 fJ so that no gain modulation due to residual 10.5-GHz pulses
appeared. Figure 2.9(c) shows examples of measured XGM waveforms. We can see
two recovery processes after gain depletion. From the slower process, we obtained
1/τC (Fig. 2.10(c)). In one exception, we assumed τC ∼ 100 ps for sample A#3
without ﬁtting, since its small gain seriously degraded the S/N ratio of the output
signal and prevented a precise determination of τC . From Fig. 2.10(c), we see that
the SOA samples can be categorized into two types: the recovery rates of B#1, B#4
and D#1 increased rapidly to 30 GHz with Iop , while those of the others increased
slowly around 5 to 10 GHz.
Using the measured fundamental parameters, we calculated the conversion eﬃciencies (Fig. 2.11). The similarity of the results for the sample B#1 and those forB#4 indicates that our measurements were accomplished precisely. The eﬃciency
values for each sample at Iop = 200 mA were η1 = 0.68 to 0.87, η2 = 0.17 to 1.30,
η3 = 0.29 to 0.53, and ηT = 0.07 to 0.40. These values indicate that total eﬃciency
ηT strongly depends on the SOA structure. This dependence resulted mostly from
η2 . The B-series MQW samples had a much larger η2 than the other bulk samples.
Among the A-series, the longer chips had a larger η2 , whereas the B-series showed
the opposite η2 dependence on Leﬀ . The reason for this diﬀerence is unclear. η125

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

(a)

(b)

SOA chip

SOA
Lensed fiber tip

Input(PM-DSF)

Output(DSF)

Bonding wires(c)
PM-DSF
SC connector

SC/PC

400~500
zirconia ferrule

15
Tapered angle 100 +/- 4 (deg)Radius of curvature 6 +/- 1.5 mm
AR coat reflectivity <0.5 %Figure 2.7: Photographs and a drawing of the experimental setup, SOA samples
and lenzed ﬁbers. (a): Lensed ﬁber coupling stages and the SOA chip mount. (b):Microscopic view of the lensed ﬁber coupling. (c): Design of the PM-DSF lensed
ﬁber.
26

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Chip Gain (dB)+40

Leff (µm)
A#1, 1000
A#2. 700
A#3, 300
A#4, 700

+30
+20
+10

B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

0
-10

20

100

500

20

100

500

C#1 700
D#1 1500

20

100

500

Nominal injection current, Iop (mA)
Figure 2.8: Measured small-signal gains of SOA samples versus current. Sample
details are given in Table 2.1. Gains were measured with cw light, λ2 = 1548.5 nm.
Input cw power into each chip was kept under −30 dBm.
depends on I0 , and decreases with Leﬀ since I0 is proportional to Leﬀ . η3 was significantly smaller than unity. The results in Fig. 2.11(c) indicate that η3 decreases as
Iop goes above the optimum condition, and that longer SOAs tend to have larger η3values at large Iop . The total eﬃciency ηT of longer SOAs also tends to be higher atlarge Iop . Sample B#1 (and B#4) will be the most economical device for Iop ∼ 400
mA (ηT ∼ 0.34), while B#3 has the best eﬃciency for Iop ∼ 150 mA (ηT ∼ 0.46).
We consider that the superiority of the MQW structure found in η2 has a diﬀerent origin from the well-known MQW superiority for LD application. The latter istypically attributed to the low internal loss (∼ 5cm−1 ) or larger diﬀerential gain at
low carrier density, but these will not explain the observed high eﬃciency at highcarrier density. High cw-saturation power of MQW SOAs have been noticed by
SOA researchers, but they do not provide convincing explanation since they do notprecisely evaluate carrier recovery rate values.
There appears to be a problem with the deﬁnition of Nex using Eqs. (2.1) and(2.2), since the calculated η2 sometimes exceeds unity. This underestimation of Nex
will be reduced if we use another deﬁnition as∫ Iop
τC (I)
Nex =
dI,
(2.6)
q
I0
taking into account that the carrier lifetime τC , which is related to the diﬀerential
carrier injection rate, decreases with the injection current. Determination of Nex

27

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Chip Gain for cw light (dB)

(a)

(b)
Chip Gain for pulse (dB)

+30

+20

+10

0

Leff = 500 µm
B#3, Iop
200 mA
150 mA
100 mA
80 mA

-30

-20

-10

0

+10 +20+30+20

Leff = 500 µm
B#3, Iop
200 mA
150 mA
100 mA
80 mA

+10

0

1

10

100 1000 10000

Output pulse energy from chip (fJ)

Output cw power from chip (dBm)(c)
Chip Gain (dB)

+40
+30

Iop
400 mA
300 mA
200 mA
150 mA+20

100 mAB#1 sample

+10
0
-200

0

+200

+400

+600

Time (ps)Figure 2.9: Typical SOA chip characteristics (a) Gain-saturation proﬁles for cw light.
(b) Gain-saturation proﬁles for ultrafast pulses (2-ps width, 0.65-GHz repetition).
The dashed lines in (b) show the theoretical ﬁt using Eq. (2.4). (c) Typical XGM
proﬁles measured with a cross correlator. The cw-probe intensity and pulse energy
into the chip were set to −25 dBm and 10 fJ, respectively.

28

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Saturation energy, Esat pulse (fJ)cw

Saturation power, Psat (mW)

(a)
100.0
50.0

Leff ( µm)
A#1, 1000
A#2, 700
A#3, 300
A#4, 700

Leff ( µm)
B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

C#1 700
D#1 1500

10.0
5.0
1.0

20

100

500

20

100

500

20

100

500

Nominal injection current, Iop (mA)(b)

3000.01000.0300.0
Leff ( µm)
A#1, 1000
A#2, 700
A#3, 300
A#4, 700

100.0
30.0

20

500

20

100

500

C#1 700
D#1 1500

20

100

500

Nominal injection current, Iop (mA)(c)
Carrier recovery rate, 1/τC (GHz)100

B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

50

Leff ( µm)
B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

A#1, 1000
A#2, 700
A#3, 300
A#4, 700

105

C#1 700
D#1 1500

1

20

100

500

20

100

500

20

100

500

Nominal injection current, Iop (mA)

Figure 2.10: Measured SOA parameters used for evaluation of the conversion eﬃciencies (a) Saturation power of SOA chips versus current for cw light. (b) Saturationenergy for ultrafast pulses (2-ps width, 0.65 GHz). (c) Carrier-recovery rate 1/τC .

29

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

η1=(Iop- I 0)/Iop

(a)
1.0
0.8
0.6
0.4
0.2
0.0

Leff (µm)
B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

Leff (µm)
A#1, 1000
A#2. 700
A#3, 300
A#4, 700

20

100

500

20

100

500

C#1 700
D#1 1500

20

100

500

Nominal injection current, Iop (mA)

η 2=Ncw/Nex(b) 1.5
1.0
0.5
0.0

(c)

η 3=Npulse /Ncw

Leff (µm)
A#1, 1000
A#2. 700
A#3, 300
A#4, 700

B#1, 1100
B#2, 700
B#3, 500

20

100

500

20

100

C#1 700
D#1 1500

B#4, 1100

500

20

100

500

Nominal injection current, Iop (mA)
1.0

Leff ( µm)
B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

Leff ( µm)
A#1, 1000
A#2. 700
A#3, 300
A#4, 700

0.8
0.6

C#1 700
D#1 1500

0.4
0.2
0.0

20

100

500

20

100

500

20

100

Nominal injection current, Iop (mA)

30

500

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Efficiency, η T = η1 η2

η3

(d)
0.6

B#1, 1100
B#2, 700
B#3, 500
B#4, 1100

Leff (µm)
A#1, 1000
A#2, 700
A#3, 300
A#4, 700

0.5
0.4

C#1 700
D#1 1500

0.3
0.2
0.1
0.0

20

100

500

20

100

500

20

100

500

Nominal injection current, Iop (mA)

Figure 2.11: Measured dependence of SOA sample conversion eﬃciency on Iop (a):
η1 , (b): η2 , (c): η3 , according to Eqs. (2.2) to (2.5), and (d): total eﬃciency
ηT = η1 × η2 × η3 .
using Eq. (2.6) is diﬃcult, however, because it requires precise measurement of τC at
small Iop . Then we think using Eqs. (2.1) and (2.2) is not a concern and appropriate
since the idea is to get a rough understanding of the SOA-characteristics.
Thus, our interpretation of the dependence of the eﬃciencies on the SOA structure can be summarized as follows. First, the MQW structure is advantageousfor increasing η2 . A longer active region decreases η1 , increases or decreases η2 , andincreases both η3 and total eﬃciency at large Iop . We could not obtain conclusive results for width, thickness or conﬁnement factor, though some of these factors seemed
to signiﬁcantly aﬀect eﬃciencies. Although these results are useful, we will need toperform more systematic characterizations before we can completely understand the
SOA-structure eﬀects.

31

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

2.4

Estimation of SOA electrical power consumption of all-optical gating

Here we propose a model for determining the electrical power requirements of SOAs
used in ultrafast all-optical gates. Our ﬁrst target is to understand the dependenceof the power requirement on the operation frequency. The other target is the power
consumption estimation based on the SOA fundamental parameters. First, we discuss the carrier dynamics of SOAs. Then, we estimate the power consumption
from the calculated carrier dynamics. Finally, we compare the calculated power
consumption with measured results.

2.4.1

Rate equation model including conversion eﬃciencies

In our preceding analyses[9, 56, 57], we used a single rate equation to calculate thecarrier-density modulation in one SOA module. This is a simple but powerful tool
to explain the gating characteristics in the time domain, as we will see in Chapter 3.
The three kinds of carrier conversion eﬃciencies had not been taken into account,however, in the previous model. To account for the eﬀect of the three diﬀerent
conversion eﬃciencies, we expanded the rate equation to the following form:()
dnpulseIop
npulsedg cw
Pcw=
η1 η2 η3 −− η3 {exp ΓLeﬀ
npulse − 1}
dt
qV
τC
dn
hνV
(
)
pulse
dg
Ppulse− {exp ΓLeﬀ
npulse − 1}
(2.7)
dn
hνV
(npulse = Npulse /V : carrier density available for ultrafast gating, Ppulse and Pcw :input light intensities, Leﬀ and V = Leﬀ wA dA : active region eﬀective length andvolume, Γ: conﬁnement factor, dg/dn: diﬀerential gain). Note that η3 is multipliedby the cw-light term, in view of the diﬀerence between npulse and ncw (the carrier
density available for cw light). Both cw gain and pulse gain were supposed to be
determined by npulse , since we did not observe much diﬀerence between them unlessgain saturation occurred (Fig. 2.9). We also supposed there are reserved carriers(nreserve = ncw −npulse ) which do not explicitly aﬀect gain, and (1−η3 ) of the photons
are converted from reserved carriers through intraband interactions in the case of cwlight ampliﬁcation. This rate-equation model can explain the experimental results
regarding the gain saturation for ultrafast pulses quite well, and those regarding
32

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Table 2.2: Estimated refractive index change of each sample, and related parameters.
Refractive index, Carrier density for Diﬀerential gain, Conﬁnement
Sample change dnr /dnpulse
gating, npulse
dgcw /dnpulse
factor,
−20
3
18
−3
−17
2
(10
cm )
(10 cm ) (*2) (10cm ) (*2)Γ (*3)
A#1
3.8
0.17
142
0.2
A#2
3.8(∗1)
0.11
196
0.2
A#3
3.8(∗1)
0.15
136
0.2
B#1
1.4
0.43
76
0.2
B#2
1.2
1.0
42
0.2
B#3
0.7
2.0
30
0.2
(*1) Estimated from the value for sample A#1.
(*2) Typical values measured at Iop = 200 mA are shown.
(*3) Assumed in view of the values in Table 2.1.
the cw-gain saturation up to 3-dB suppression points (Fig. 2.12(a)). For stronger
cw light, other theories may provide better predictions of the gain suppression[58].
We also mention that the sub-pulse intensities calculated using this model showed
less agreement with measured results, than those described in Chapter 3. Thus welimit the use of this model within this chapter, until more comprehensive model isdevised.
The rate equation can be used to calculate the optical carrier modulation andnonlinear phase shift. Injection of Ppulse causes carrier recombination ∆npulse , andthe nonlinear phase shift is given byαdg∆Φ = − ΓLeﬀ ∆npulse = k0 dnr /dnΓLeﬀ ∆npulse
2
dn

(2.8)

(α: α parameter, k0 : wave number in vacuum, nr : refractive index).
To verify the model, we simulated the SOA characteristics under holding-beam
injection. We used the measured values of η and τC at each Iop , and extracted
dgcw /dn and dgpulse /dn from the corresponding SSG values. Figure 2.12(b) shows
measured and calculated results for the eﬀective carrier recovery rate of sample B#3
accelerated by the holding-beam. Figure 2.12(c) shows measured and calculated
results for pulse-gain saturation with and without a holding beam. We see thatboth results approximately agree with each other.
We also compared measured and calculated nonlinear phase shifts for samples
A#1 and B#1 to #3. These can be experimentally determined from XPM spectraout of the SOA[9]. The chosen pulse frequency was either 2.6 GHz or 10.5 GHz,
33

Chip Gain (dB)

+40
+30
+20
B#1, Iop = 100, 150,
200, 300, 400 mA
measured
calculated

+10
0

-20

-10

0

+10

+20

Recovery Rate 1/τeff (GHz)CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

105-50 -40 -30 -20 -10 0+30with -5 dBm holding beam

+20

B#1, IOP = 400 mA

1
0.5

0.2

+10
0

holding beam intensity
measured calculated-25 dBm-15 dBm-5 dBm0.1

measured calculated
Iop
400 mA,
200 mA,

0.05
0.1

1
10
100
1000 10000
Output pulse energy from chip (fJ)

+10 +20 +30

Holding-Beam Intensity (dBm)2

B#1 sample∆Φ/π (rad)Chip Gain (dB)without holding beammeasured
calculated

50Output power (dBm)+40

B#3, Iop = 200 mA

100

1

10

100

1000

Pulse energy (fJ)

Figure 2.12: Comparison of measured and calculated SOA properties under holdingbeam injection. (a) Cw-gain saturation, (b) eﬀective carrier recovery rate 1/τeﬀ , (c)gain saturation for a 2-ps pulse, and (d) nonlinear phase shift ∆Φ caused by ultrafast
pulses.

34

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

depending on the carrier recovery rate of the SOA. dnr /dn was estimated for eachsample from the measured phase shifts at Iop = 200 mA (Table 2.2). The diﬀerence
between samples seemed to be negatively correlated to the typical carrier densities.
The SOA simulation could approximately reproduce the measured nonlinear phaseshifts at various Iop , Pcw and Ppulse values (Fig. 2.12(d)), and our model seems
feasible for power-consumption simulations.

2.4.2

Electrical power consumption

We calculated the correspondence between the electrical power consumption Pop
and the operating frequency B for an SOA used in an XPM-based all-optical gate∫∞
as follows. First, we ﬁxed the control pulse energy Epulse ≡ −∞ Ppulse (t)dt at arealistic value (340 fJ). Then, for each Iop condition we calculated the induced ∆Φ asa function of the holding beam intensity. The required amount of ∆Φ depends on thefunction of the gate: for 3R regeneration or wavelength conversion it is ∼ 0.3π[56].
Therefore, for each Iop , the equation ∆Φ = 0.3π provides the maximum available
holding-beam intensity Pcw . The carrier recovery rate 1/τeﬀ for this condition canbe interpreted as the approximate frequency limit B of the gate. The actual B fora speciﬁc gate may depend also on other factors, as BPF detuning in Chapter 4,
but we do not discuss such detail in this chapter. Pop was calculated from Iop using
empirical V -I characteristics.
Before discussing the power consumption of actual SOAs, we will look at calculated values using non-empirical parameter sets to ﬁnd their dominant factors.
Figure 2.13 shows our results. Here, the dependences of dg/dn, η and τC on Iop
were ignored, in contrast to the cases shown in Figs. 2.12 or 2.14. Calculations
were done down to nearly the low power limits, where ∆Φ = 0.3π can be satisﬁed
only when Pcw = 0. We found that the maximum frequency is almost proportional
to the total conversion eﬃciency ηT and dg/dn, but not sensitive to the intrinsic
carrier recovery rate 1/τC . As for the conversion eﬃciencies of each process, η3 affects the required holding-beam intensity as well as B. As long as we keep the SOA
parameters constant, Iop increases almost linearly with B, so Pop increases with B 2 .
Figure 2.14 shows the calculated power consumption for the SOA samples. We
used the SOA parameters from §2.4.2 . The B-series shows better performance thanthe A-series. When Pop is small, B#3 has the best performance around 20 GHz.
As the target frequency increases to over 100 GHz, B#1 becomes preferable. The
power consumption of B#1 at 160 GHz is about 750 mW. This is comparable to the35

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATESPOP = VOP IOP (mW)at ∆Φ=0.3 π

30000

3000

η T=1, τC =100 ps, dg/dn=20
τC = 50 ps
τC =200 ps,
η T=0.5-17
dg/dn= 40 10-17
dg/dn= 10 10

-17

2

10

cm

Demux,(Electrical)ref [45]300
2(1/τeff)
30
1

10

100

1000

Bandwidth, 1/τeff (GHz)
Figure 2.13: Calculated power consumption of SOAs versus their maximum operating frequencies when SOA parameters are independent of Iop (Table 2.3).

Table 2.3: Parameters assumed for Fig. 2.13
SymbolDescription
Value
Leﬀ
active region length
1000 µm
wA
active region width
1 µm
dA
active region thickness
0.05 µmΓ
conﬁnement factor
0.2
dgcw /dn
diﬀerential gain for cw
20.0 × 10−17 cm2
dgpulse /dn
diﬀerential gain for pulse
20.0 × 10−17 cm2
τccarrier recovery rate
100 pswithout acceleration
drr /dnpulse
refractive index change
0.4 × 10−20 cm3
ηT
total carrier-conversion eﬃciency
1.036

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES(a)POP = VOP I OP (mW)at ∆Φ=0.3 π

3000
1000

B#1 1100 µm
B#2 700 µm
B#3 500 µm
A#1 1000 µm
A#2 700 µm

300
100
30

3

(b)
3000
POP = VOP I OP (mW)calculated

2(1/τeff)

1030
100 300
Bandwidth, 1/τeff (GHz)measuredB#4 1100 µm
B#3 500 µm

2(1/τ eff )

1000
300
100
30
3

calculated

B#1 1100 µm
B#3 500 µm

1030
100 300
Bandwidth, 1/τeff (GHz)

Figure 2.14: SOA electrical-power consumption versus carrier recovery rate underholding-beam acceleration. (a): Calculated results using the measured parameters
for each sample. (b): Measured results for sample B#3 and B#4. A control pulse
with a 2-ps width and 340-fJ energy was used.

37

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

power consumption, about 400 mW, of the latest XPM-based 160-Gb/s wavelength
conversion[23] or wavelength conversion using ultrafast chirp dynamics[25]. For the
A-series, the longer sample had a higher frequency, and the shortest sample A#3
could not achieve ∆Φ = 0.3π under the injection-current limit. These characteristics
result from those of ηT as discussed in §2.3 . Except for the diﬀerence in scale, mostof the Pop -B proﬁles in Fig. 2.14 resemble the corresponding Pop -1/τC proﬁles. Thisis due to the complicated correlations of SOA parameter changes. Consequently,
Pop of A#1 and B#1 increase with B 2 within the range of about 300 to 1000 mW,
and the corresponding values of the others likely increase faster than that. Further
discussion requires measurement of the eﬃciencies at larger Pop .
To test the validity of our power-consumption model, we measured Pop and 1/τeﬀ
at ∆Φ = 0.3π for sample B#3 (Fig. 2.14). We obtained good agreement between themeasured and calculated results within a frequency range of 20 ∼ 40 GHz. We also
tried to measure that for sample B#4 and compared with the calculated results forB#1. In this case, we could only reduce ∆Φ down to ∼ 0.38π even with our strongest
holding beam (+11 dBm into chip). Thus the each plot for this series indicates only
lower limit of the bandwidth at each power consumption. For this SOA, somewhat
larger discrepancy appeared between measured and calculated results. Thus weconsider that our model has to be improved to provide absolute value estimationin the frequency range around and over 100 GHz. As an initial power-consumption
model, nevertheless, we believe that our model is valuable enough. Especially, theB 2 law has become more convincing due to the measured results.

2.4.3

High-frequency limit

To discuss the future of photonic networks, it is useful to express the B-Pop relation
with a simple formula. In this section, we therefore derive an analytical expression
for the high-frequency limit.
First, we assume that both the injection current Iop and holding-beam power
Pcw are large. Pcw is chosen according to Iop so that ∆Φ stays at a ﬁxed value,as in §2.4.2. Without control pulses the carrier density reaches an equilibrium at
npulse = neq . The small diﬀerence in dgcw /dn and dgpulse /dn is ignored for simplicity,and the saturated SOA gain is expressed as)dg
Geq = exp ΓLeﬀ neq .
dn
(

(2.9)

38

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Note that we can approximately suppose that Geq does not change with Iop , becauseof the constraint on ∆Φ. The reason is as follows: The number of carrier-to-photon
conversions when a control pulse is injected into the SOA in the steady state is given
from the net gain of the pulse Gpe and Epulse :∆npulse ≈Gpe − 1
EpulsehνV(2.10)

as long as the pulse width is short enough compared to the carrier relaxation time.
Then, Gpe can be roughly estimated from Geq . Since ∆Φ is given from ∆npulse by
Eq. (2.8), we can transform Eq. (2.10) into
Geq ≈ Gpe ≈ 1 +hνV ∆Φ
.
Epulse ΓLeﬀ k0 dnr /dn(2.11)∗)

Both ∆Φ and Epulse are regarded as constant when we obtain the B-Pop relation.
We also assume that dnr /dn does not depend on Iop . Therefore, we can approximately regard Geq as constant. For example, ∆Φ = 0.3π and Epulse = 340 fJ resultin Gpe = 2.5 for sample B#1.
By assuming a steady state and taking only terms including Iop or Pcw in Eq.
(2.7), we obtain the following relation from Eqs. (2.7) and (2.9):

Pcw
ηT Iop≈hνVη3 (Geq − 1) qV(2.12)The operating frequency limit can then be obtained as follows: After ﬁrst-order
expansion of Eq. (2.7) in terms of npulse around neq , the relaxation of the carrier
density deviation ∆n(t) = npulse (t) − neq after the control-pulse injection can be
expressed asd (∆n (t))∆nGeq ΓLeﬀ dg=−− η3
Pcw ∆ndt
τChνV dn= −B · ∆n(2.13)

After substituting Eq. (2.12) into the above, we obtain
B≈∗)

Geq ΓLeﬀ dg
1+(Iop ) · ηT (Iop ) · Iop .
τC (Iop ) Geq − 1 qV dn

Note that this is a fairly rough approximation.

39(2.14)

CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

When 1/τC converges to a ﬁnite value as Iop increases, the second term will dominate.
The applied voltage Vop approaches RIop (R: resistance). Then, when we assume
both dg/dn and ηT are independent of Iop , Pop is expressed as(

Geq − 1 qV
Pop ≈ R
Geq ΓLeﬀ= constant × B 2 .

)2 (

dg
dn

)−2

ηT−2 B 2(2.15)

Therefore, the B-Pop relation will be quadratic under the above assumption.
This explains the Fig. 2.13 results. The total conversion eﬃciency has an inversesquare contribution to Pop for the actual SOA, so it should be elaborately optimized
to reduce the dc power consumption.
In a limiting case when we can use arbitrary-high energy control pulses, we canobtain even simpliﬁed expression of the operation frequency limit. In this case,∆npulse is equal to neq . Using this relation and Eq. (2.8), we obtain
neq =

2∆Φ
.
αΓLeﬀ dndg
pulse

(2.16)

From Eq. (2.9) and Eq. (2.16), we obtain
Geq = exp2∆Φα(2.17)

instead of Eq. (2.11). The derivation of Eq. (2.15) is still valid, and after substituting Eq. (2.17) into Eq. (2.15) we obtain
B2
Pop ≈
,
M
((
(
))
)−2 ( )2
2∆Φ
dA wA
dg
−1
M ≡R
ηT2
1 − exp −qαΓ
dn(2.18)

The factor M in this equation depends only on the SOA parameters (except for∆Φ = 0.3π), and we can regard this quantity as a ﬁgure of merit of the SOA in
terms of the power consumption. Table 2.4 shows M values for the measured SOA
samples. For simplicity, the dependence of each parameter values on the injection
current was ignored. This ﬁgure can be used for crude comparison of SOA samples,though it does not properly reﬂect decrease in the eﬃciency at large injection current.

40CHAPTER 2. FREQUENCY-DEPENDENT ELECTRIC POWER CONSUMPTIONMODEL INCLUDING QUANTUM-CONVERSION EFFICIENCIES IN ULTRAFASTSOA GATES

Table 2.4: Figure of merit values of the SOA samples
α parameter,
Total
Diﬀerential gain,
Figure ofSampleαconversion eﬃciency
dgcw /dnpulse
Merit,M
−17
2
2
ηT
(10
cm )
(10 GHz2 /mW)
A#1
2.2
0.15
142
0.29
A#2
1.5
0.11
196
0.15
A#3
2.3
0.084
136
0.03
B#1
1.5
0.34
76
2.43
B#2
2.3
0.41
42
1.88
B#3
1.9
0.34
30
0.45

2.5

Conclusion

We developed a new model of the electrical power consumption of SOAs used inultrafast all-optical gates. The dominant factors of this model are three diﬀerent
carrier-conversion eﬃciencies and the diﬀerential gain of the SOA.
To predict power consumption, we measured the conversion eﬃciencies of current
SOAs with our original method. They were ηT = 0.07 to 0.46 in total. A systematic
study of the eﬃciencies of SOAs with diﬀerent structures revealed that a longer
SOA with an MQW structure will have high conversion eﬃciency at a high injection
current. For high-frequency operation above 100 GHz, we consider that the SOA
eﬀective length of more than 1100 µm will be preferable. Optimum SOA-length will
exist depending on the target frequency, which should be clariﬁed by further study.
Also further study is needed with this method to understand the dependences on the
SOA width, thickness, and conﬁnement factor, and through such studies we hope
to design a more eﬃcient SOA in the near future.
We calculated the power consumption Pop as a function of operating frequency
limit B, based on our power consumption model and measured SOA parameters.
We predicted that the dc power consumption of SOA gates Pop increases with theiroperation frequency B as B 2 , or even faster, depending on the decrease in theconversion eﬃciencies and diﬀerential gain. Measured results for two SOA samples
on the power consumption showed clear B 2 dependence within the frequency rangeof 20 ∼ 100 GHz. Our ﬁrst power-consumption model will be a fundamental fordeveloping the power-consumption models in in the frequency range above 100 GHz.

41

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Chapter 3
Observation of sub-pulse generation from the
DISC-type wavelength converter

3.1

Introduction

As we introduced in Chapter 1, several types of all-optical SOA gates have been
intensively studied for use in future OTDM-WDM networks and systems. Notably,
DISC and its variants have led to some of the most successful achievements in thelast decade. 168 Gb/s error-free wavelength conversion [23] and 40-Gb/s error-freeultralong-distance 3R transmission [29] have been demonstrated using DISCs intheir original conﬁguration, and a 320 Gb/s wavelength conversion has also been
demonstrated using transient cross-phase modulation in the SOA[26].
It has been noticed, however, that the eye diagram of the DISC output signal
often suﬀers from serious degradation, which could not be ascribed to the wellknown pattern-induced intensity ﬂuctuation (PIF). Some modeling research studies[57] have been focused on explaining this phenomenon, and one fundamental issue
has been recognized as the potential source: the generation of small sub-pulses between RZ-formatted output pulses.
In this chapter, we explain our ﬁrst observation of sub-pulse generation througha gate experiment and its theoretical analysis. In addition, we discuss measured
results on the theoretically predicted increase in the sub-pulse intensity with thecarrier recovery rate of the SOA. From the measured and calculated results, weconclude that our model is applicable for use in a wide recovery rate range. Finally,
we discuss the trade-oﬀ between the sub-pulses and PIF associated with the recovery
rate change in a wide range. This result illustrates that sub-pulse generation can bea signiﬁcant limiting factor for practical DISC operations.
42CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

3.2

Sub-pulse generation modelTwo phenomena have been suspected as potential sources of sub-pulse generation.
One is the exponential recovery proﬁle of the SOA carrier, and the other is theultrafast SOA response due to the carrier-heating eﬀect. In this section, we explain
the sub-pulse generation of the ﬁrst kind using our DISC-gate model. Generationof the latter sub-pulses is beyond the scope of this research.

3.2.1

DISC-gate Operation model based on the SOA Carrier Dynamics

A DISC gate in its original concept consists of a single SOA as an optical nonlinear
element, and a Mach-Zehnder Interferometer (MZI), which determines the output
timing window (Fig. 3.1(a)). The input pulse train with a wavelength of λ1 iscombined with continuous-wave (cw) light (λ2 ) and injected into the SOA. The alloptical modulation process can be modeled based on the rate equation of the excess
carrier density in the SOA as]) [ |E2d
ηT Iop nc (t)1 (|Ecw |2
pulse (t)|nc (t) =−−G(nc (t)) − 1
+
.
dt
qV
τC
V
hνpulse
hνcw

(3.1)

Here, nc (t) is the average nc (z, t) over the active region length. nc (z, t), the carrier
density which contributes to the SOA gain, is diminished from the total carrier
density, as represented by the quantum conversion eﬃciency ηT . Iop is the injection
current, τC is the carrier lifetime, V is the active region volume, and Epulse (t) and
Ecw are the amplitudes of the input optical signals. This model is almost same asthat used in the past analyses[9, 56, 57], except for one point that the total carrier
conversion eﬃciency is explicitly included. Meanwhile, the three diﬀerent conversion
eﬃciencies which we introduced in §2.2.1 is not included at the moment. The ﬁrst,
second, and third terms on the right-hand side of eq. (3.1) represent the carrier
injection, carrier relaxation, and stimulated recombinations, respectively. Epulse (t)for a clock operation is expressed as
Epulse (t) =

∞
∑
m=1

√

Epulsesech
2TW(t − mT
TW

43

)
,

(3.2)

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

(a)
∆ΦB

IOP
data input
λ1
cw

SOA

(c)

[x1016]

∆Φ(t)

4
3
2
1
0

-100

λ2

[π]

+0.0

δ∆Φ(t)

Excess carrier density
nc(t) (1/cm3 )

5

BPF

∆t

λ2

(b)

λ2
data output

0

+100

+200

+300

Time (ps)

∆t

-1.0

∆Φ(t)+∆ΦB∆Φ(t-∆t)

-2.0
+1.5
+1.0
+0.5

-25

0

+25

+50

Time (ps)Figure 3.1: The schematic and operation principle of the DISC gate: (a): Schematicof the gate. (b): Calculated example of the time proﬁle of the SOA carrier density
during optical modulation. (c): Time proﬁles of the nonlinear phase shifts for the
split optical components in the MZI, and the phase diﬀerence between them.

44

+75

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

where Epulse is the pulse energy and TW is a parameter related to the full-width at
half-maximum pulse width TFWHM as
(

(

√ ))
TW = TFWHM / 2 ln 1 + 2 .

(3.3)

G, the chip gain of the SOA, is expressed as()
dg
G(nc (t)) = exp ΓLeﬀ nc (t) ,
dn

(3.4)

where Γ is the conﬁnement factor, Leﬀ is the eﬀective length of the active region, anddg
is the diﬀerential gain of the active region. nc (t) decreases through stimulateddnrecombinations as each control pulse incidents on the SOA (Fig. 3.1(b)), and thisdecrease causes an ultrafast change in the refractive index nr of the SOA throughthe band-ﬁlling eﬀect. Then the co-propagating cw signal suﬀers nonlinear phasemodulation expressed as∆Φ(t) = k0 ΓLeﬀ

dnr
∆nc (t).
dn(3.5)

Here, k0 = 2π νcw
is the wave number of the light in vacuum. ∆Φ(t) increases withincless than one picosecond after each pulse injection, but recovers slowly according to
the eﬀective carrier relaxation time τeﬀ . The delayed interference scheme has been
adopted to inhibit this slow response as follows: The gain-modulated and phasemodulated cw light is split into two components at the MZI, which are combined
after suﬀering a time diﬀerence ∆t and a phase diﬀerence (phase bias) ∆ΦB at theend of the MZI. Then the amplitude of the combined signal is expressed as√
1 √Eout (t) = { G(t) exp [i (∆Φ(t) + ∆ΦB )] + G(t − ∆t) exp [i∆Φ(t − ∆t)]}Ecw ,2(3.6)
and the output signal intensity Pout (t) is expressed as
[
]
√
1
2
2
Pout (t) = |Eout (t)| =G(t)G(t − ∆t) cos (δ∆Φ(t)/2) + δG(t) |Ecw |2
4
(3.7)
2

δ∆Φ(t) ≡ ∆Φ(t) − ∆Φ(t − ∆t) + ∆ΦB√√δG(t) ≡ G(t) − G(t − ∆t)

45

(3.8)
(3.9)

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Thus, the output signal intensity is dominated by the phase diﬀerence δ∆Φ(t). It islargely shifted from ∆ΦB only within t0 < t < t0 + ∆t, where t0 is the pulse incident
time, and the output gate window is shortened to ∆t (Fig. 3.1(c)). Since the input
data pulses at λ1 do not pass the band-pass ﬁlter (BPF), we obtain wavelengthconverted signals from the DISC gate.

3.2.2

Sub-pulse generation model

There remains a problem with this scheme, however, since the recovery of ∆Φ(t)is not linear-function-like but exponential with t expressed as exp(−t/τeﬀ ). Figures
3.2(a) to 3.2(d) show exaggerated examples of the phase modulation proﬁles andand the output waveform. Since the degree of the phase modulation for the twocomponents ∆Φ(t) and ∆Φ(t − ∆t) at a certain time position is not equal, the phase
diﬀerence δ∆Φ temporarily goes below ∆ΦB after t > t0 +∆t. Transient cancellationis possible if we properly detune the ∆ΦB from π, but complete cancellation of thephase shift after t > t0 + ∆t is not possible since d∆Φ(t)and d∆Φ(t−∆t)
do not balancedt
dt
as it is. Therefore there occurs a sub-pulse outside the gate window, potentiallycausing a bit error. Note that if we can induce asymmetric phase shifts for thetwo components as φ1 ∼ exp(−t/τeﬀ ) and φ2 ∼ fphase · exp(−(t − ∆t)/τeﬀ ), we cansuppress sub-pulse generation by selecting fphase = exp(−∆t/τeﬀ ). As we will showin §3.4.2, this sub-pulse generation becomes a serious issue as we increase the SOA
carrier recovery rate compared to ∆t and so on.

3.3

Experimental Setup for Sub-pulse Measurement

3.3.1

Overview

The generation of sub-pulses had not been revealed clearly before our measurement
owing to the lack of monitoring resolution in previous experiments. Precise, quantitative investigation of sub-pulses requires waveform monitoring in a high dynamic
range with a high time resolution over a long time span. Thus, we employed a crosscorrelator (Femtochrome Research Inc., FR-103XR) that had a dynamic range exceeding 27 dB. Its time resolution was determined by the width of optical sampling
pulses, and 2.2 ps could be achieved using pulse trains from a frequency-tunable
mode-locked ﬁber laser (MLFL: Pritel Inc., UOC-3).
46

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER(π rad)

(a)

Φ(t)

0.0
+1.5 (b)cos2(δΦ(t)/2)

Φ(t-∆t)

(d)

δΦ(t)

(π rad)1.0(c)∆T=5ps,τeff=5 ps

0.0
50
Pout(t)

(mW)

+1.0

+1.0

0
0

Time (ps)40 0Time (ps)40

Figure 3.2: Time proﬁles of the (a) nonlinear phase shifts, (b) and (c) phase diﬀerence, and (d) gate output signal when τeﬀ is short.

Table 3.1: Measured characteristics of the SOA#10 used in §3.4.1 at Iop = 250 mA.
Parameter
G0
sat
Pcw
sat
EpulseτC

Description
small signal gain
coupling loss against pigtail ﬁbers
saturation power for cw lightsaturation energy for 2-ps pulse
carrier lifetime

47

Value
23 dB
2 dB
14 mW
600 fJ
200 psCHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER(a)
DISC gate

VOA P H Q
CW light (λ2) EDFAMZIDFB LD∆t=5.0 ps
calcite

100m SMFQEDFAVOA P H

output pulses (λ2)

SOA #10

BPF
3nm

MLFL

isolator Q1

MUX

Q 2 P2

P1 C1

12.5GHz

input pulses (λ1)
EDFA

BPF

probe

Q H

input 5nm
cross
correlator

digital storage
oscilloscope

(b)

EDFA

HQ
spectrometer

DISC gate
Power
Meter

Power
Meter

circulator
Power
Meter

PBS

PM-DSFlensed lensed
fiber
fiber
SOA
chip B#1

MZI∆t=5.0 ps
calciteQ1 P1 C1

BPF
3nm

Q 2 P2

FP-LD (λassist)
Figure 3.3: Setup of the DISC-gate experiment. (a): Setup for §3.4.1. VOA: variable
optical attenuator, MZI: Mach-Zehnder interferometer, SOA: semiconductor opticalampliﬁer, Q: quarter-wave plate, H: half-wave plate, P: polarizer, MLFL: Modelocked ﬁber laser, MUX: optical time-division multiplexer. (b): Modiﬁed DISC-gate
setup for §3.4.2.

48

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

(c)

+40

+40

+20

+20
0

0

SOA #10
SOA B#1

-20
100

30

300

1000

Nominal injection current (mA)

(b)
+40

Chip Gain (dB)

25 GHzcw 5 mW∆ΦNL~ 0.54 πpulse 475 fJ(chip)
SOA #10,+30+20
SOA #10, 250 mA
SOA B#1, 400 mA

+10
1

10

100

-20
-40
-60
+40

(d)

0
-20

-60
1547

1000 10000

cw 0.3 mW
pulse 30 fJ(chip)+20-40

2 ps, 0.65 GHz, 1555 nm
0Intensity (dBm/nm)

Chip Gain (dB)(a)∆ΦNL~ 0.69 π

SOA #2, 10.5 GHz
1548

1549

Wavelength (nm)Output energy from chip (fJ)

Figure 3.4: Measured characteristics of the SOAs. (a): Small-signal gains for cw
light, λ = 1548.5 nm. (b): Gain saturation proﬁles for temporally-isolated pulses.
Dashed lines show ﬁt results with a theoretical function. (c) and (d): Spectra of
the cross-phase-modulated cw outputs from the SOAs. Dashed lines indicate thecw-light wavelength.

49

1550

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

The setup for the waveform monitoring in §3.4.1 is shown in Fig. 3.3(a). In
this case, we used a commercially available SOA module #10 (InPhenix, Inc., IPSAD1503, drive current = 250 mA) in the DISC. Some of the measured characteristics of this SOA are shown in Table 3.1 and Figs. 3.4(a) and 3.4(b). Particularly,
precise characterization of the gain saturation for temporally isolated optical pulses
was indispensable for the accurate sub-pulse calculation in §3.4.1. In addition, thedegree of nonlinear phase shift for particular optical inputs was estimated from ther
XPM spectrum, as shown in Fig. 3.4(c), to determine dnlater. We used a set
dnof high-precision, high-extinction-ratio polarization-control devices Q1 ∼ P2 (Optoquest Inc.) in order to construct the MZI. The rotating quarter-wave plate Q1
and the rotating polarizer P1 were used to adjust the polarization of the cw lightto exactly 45o oﬀ the principal axis of the calcite crystal C1 (DGD, ∆t = 5 ps).
The next quarter-wave plate Q2 and the rotating polarizer P2 were used to control∆ΦB precisely. Details of this polarization-based MZI will be discussed in §3.3.2.
25 GHz and 12.5 GHz clock pulses were generated from the MLFL and an optical multiplexer to drive the DISC gate. The width of the pulses sent to the DISC
was broadened from 2.2 to 3.8 ps with a 100-m-long single-mode ﬁber, to suppress
carrier-heating-related phenomena inside the SOA[49–53]. The properties of the
12.5 GHz pulses are shown in Figs. 3.5(a)-3.5(c). The pedestals of the input pulses
were minimized to the best, and we obtained good extinction ratio between the RZ
input pulses (∼ 30 dB). The polarization of cw light from the distributed-feedback
laser diode (DFB-LD) as well as the clock signal was aligned to either the TE or
TM axis of the SOA at the SOA input facet, to suppress the nonlinear polarization
rotation[63, 64].
For the measurement discussed in §3.4.2, where a much faster SOA was required,
we used the custom-designed SOA chip B#1 (see §2.3) instead of the previous SOA
module. The small-signal gain, the gain saturation proﬁle, and the example of
XPM spectrum are shown in Figs. 3.4(a), 3.4(b), and 3.4(d), respectively. Part ofthe experimental setup was modiﬁed as shown in Fig. 3.3(b). Signal injection to the
SOA chip was achieved using the PM-DSF lensed ﬁber as in §2.3, with its PM axis
aligned to the TM axis of the SOA. Counter-propagating assist light of λassist = 1480
nm was used in some measurements to obtain an eﬀective recovery time as short asτeﬀ = 11 ps. The polarization-based MZI was improved so that all the components
were directly connected to each other without ﬁber connections (Fig. 3.6). This
reduced the drift of the polarization-transfer property inside the MZI.

50

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Auto correlation

1000
300
100
30
10
3
1
0.3
0.10

50

150

Time (ps)(c)
Intensity (dBm/nm)

100(b) DISC inpout, cross correlation
Signal intensity (arb.unit)

Signal intensity (arb.unit)

(a)

+20
+10

1000
300
100
30
10
3
1
0.3
0.10

50

100

Time (ps)res = 0.2 nm

λ2

0

λ1

-10
-20
-30
-40
1540

1550

1560

1570

Wavelength (nm)Figure 3.5: Measured properties of the 12.5-GHz input pulses. (a) Auto-correlation
trace (solid) of the 2.2-ps pulses from the MLFL. Dashed curve shows a deconvoluted
waveform (two-component sech-pulses). (b) Cross-Correlation trace of the DISC
input (3.8 ps). (c) Spectrum of the input pulses and cw light.

input

Q1 (H1) P1 C1 Q2 P2

outputFigure 3.6: Improved polarization-based MZI51

150

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

3.3.2

Phase bias of polarization-based MZI

As we discussed in §3.2, the MZI in the DISC plays a critical role in wavelength
conversion and sub-pulse generation. Since the sub-pulse waveform is quite sensitive
to the MZI phase bias as we will show in §3.4.1, precise determination of the bias
value ∆ΦB is essential for comparison of the measured and calculated results. In thissubsection, therefore, we discuss how the set of wave plates, calcite and polarizers
acts as an MZI, how we can determine ∆ΦB from the wave-plate settings, and theprecision of ∆ΦB .
In the ﬁrst step, cw light is evenly split into two orthogonally polarized components aligned to the C1 axes, by adjusting the Q1 , P1 , and C1 angles. Seen from oneof the C1 axes, the Jones vectors of these components are expressed as(1
0

Ein1 =)(, Ein2 =0
1)(3.10).

The Jones matrices of a calcite with an optical path length diﬀerence of ∆l, aquarter-wave plate, a polarizer, and a rotation operation are expressed as
(

)
(
)
(
)
( π)
exp 2πi ∆l
0
exp
i
0
λ
4
(
) ,
MC =
, MQ =
0
1
0
exp −i π4
(
)
(
)
1 0
cos (θ) − sin (θ)
MP =,R(θ) =,0 0
sin (θ) cos (θ)(3.11)

respectively.
In the MZI used in §3.4.1, a dispersion-shifted single-mode ﬁber was used toconnect C1 and Q2 . The Jones matrix of a single-mode ﬁber is generally expressedby a unitary matrix[61] as(MF =cos (θM ) exp (iφM )sin (θM ) exp (iψM )− sin (θM ) exp (−iψM ) cos (θM ) exp (−iφM )),(3.12)

where θM , φM , and ψM are parameters to be measured separately.
Thus, the two components Ein1 and Ein2 are converted toEout1 = MP R (− (θP2 − θQ2 )) MQ R (−θQ2 ) MF R (θC1 ) MC Ein1 ,Eout2 = MP R (− (θP2 − θQ2 )) MQ R (−θQ2 ) MF R (θC1 ) MC Ein2 ,52(3.13)

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

where θC1 , θQ2 , and θP2 are the angles of the corresponding devices’ axes. Thenthe complex ratio of these parallel vectors determines ∆ΦB (Q2 , P2 ) and the real
amplitude ratio r (Q2 , P2 ) as
Eout2 = Eout1 × r exp (i∆ΦB ) .

(3.14)The output signal of the MZI is given by Eout = Eout1 + Eout2 . We call r the
mixing ratio. Eout becomes zero when r = 1 and ∆ΦB = π. This can be achieved byadjusting θQ2 and θP2 properly, so that Q2 eliminates the ellipticity of the combined
signal and P2 extinguishes its outcome. Slight detunings of θQ2 and θP2 from these
conditions lead to a shift in ∆ΦB . Figure 3.7 shows the calculated dependence of∆ΦB for the MZI used in §3.4.1 on ∆θQ2 , while keeping ∆θP2 − ∆θQ2 constant(∆l/λ = −973.392, θC1 = 68o , θM ∼ 0.85 (rad), φM ∼ 1.98 (rad), ψM ∼ 2.76 (rad),θQ2 ∼ 131o +∆θQ2 , θP2 ∼ −64.5o +∆θP2 ) Keeping ∆θP2 − ∆θQ2 constant was an
approximate means of reducing r shift, and it was shifted from 0 to +2o in order
to slightly compensate for the r shift at around ∆θQ2 = −3 ∼ −6o . Then ∆ΦBd∆ΦB= −7.3 × 10−3 π (rad/deg). Thus, we could
shifted linearly with ∆θQ2 , and d∆θQ2accurately control ∆ΦB (∼ ±0.004π (rad)) through precise (∼ ±0.5o ) adjustmentsof wave-plate angles. Here we note that the actual ﬁber transfer matrix MF slightly
drifted with time. To compensate for this issue, we periodically adjusted θQ2 andθP2 to angles where ∆ΦB became π in the case in §3.4.1, and carefully compensatedfor the angle drifts.

3.4

Results

3.4.1

Sub-pulse measurement and calculation with systematic MZI-bias shifts

In this subsection, we explain our ﬁrst observation of sub-pulses. First, we operated
the DISC at 25-GHz operation frequency, with clock pulse input (+12.3 dBm, λ1 =1560 nm) and cw input (+9 dBm, λ2 = 1548.5 nm) into the SOA module #10,
and observed its output waveform. Figure 3.8(a) shows a typical cross-correlation
trace of the DISC output that we measured after maximizing the extinction ratio
by adjusting Q2 and P2 . As shown in Fig. 3.8(a), the extinction ratio was limited
to 25 dB by small sub-pulse-like components between the output pulses. Then, toobserve the waveform of the remaining components in more detail, we dropped the53CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

2.0

+1.10

+1.05
1.0
+1.00

+0.95

+5

0

-5

mixing ratio r∆ΦB/π (rad)∆θP2 -∆θQ2 = +2.0 (deg)

0.0
-10

∆θQ2 (deg)Figure 3.7: Calculated dependence of the MZI-bias (solid) and the mixing ratio(dashed) on the quarter-wave plate angle.
pulse frequency from 25 GHz to 12.5 GHz and carefully optimized the wave plateangles once again. The average power of the 12.5 GHz pulse train was +13.5 dBm.
At this frequency, the extinction ratio was limited to about 18 dB by several typesof relatively large sub-pulse-like components depending on the Q2 and P2 settings(Figs. 3.8(b) - 3.8(d)).
After that, we calculated the DISC output waveform to demonstrate that theabove sub-pulse-like components actually result from the operating principle modeled in §3.2. First, we chose the parameter values for use in the simulation, taking
measured chip gain, saturation energy, and the gain recovery time of the SOA into
account (Table 3.1). The saturation energy was determined to be 600 fJ from therpulse-gain saturation measurement (Fig. 3.4(b)). dn
was chosen so that the amountdnof the cross-phase-modulation in the 25 GHz operation becomes 0.5π (Fig. 3.4(c)).
The chosen parameter values are summarized in Tables 3.2 and 3.3. Then we calculated output waveforms using eqs. (3.1)∼(3.9). For the numerical integration of
the diﬀerential equation eq. (3.1), we employed second-order Runge-Kutta method.
Fig. 3.9 shows the ﬂow chart of our waveform simulation, including long-bit-pattern
waveform handling scheme to calculate pattern-induced intensity ﬂuctuation in thelater section.
We could obtain sub-pulse waveforms quite similar to the measured data, as
54

(a)

(b)

Signal intensity (arb.unit)

Signal intensity (arb.unit)

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

50

Signal intensity (arb.unit)

150

Time (ps)

(c)
1000
300
100
30
10
3
1
0.3
0.10

100

∆ΦB =1.058π

50

100

150

Time (ps)Signal intensity (arb.unit)

1000
300
100
30
10
3
1
0.3
0.10

1000
300
100
30
10
3
1
0.3
0.10

(d)
1000
300
100
30
10
3
1
0.3
0.10

∆ΦB =1.036π

t0

50

100

∆ΦB =1.072π

50

100

Time (ps)Figure 3.8: Measured cross-correlation traces of the DISC output with diﬀerent
Q2 and P2 settings: (a) with +12.3-dBm, 25-GHz input pulses, (b)-(d) with +13.5dBm, 12.5-GHz input pulses ((b)∆θQ2 = −3 (deg), ∆θP2 = −1 (deg), (c)∆θQ2 = −6(deg), ∆θP2 = −4 (deg), (d)∆θQ2 = −8 (deg), ∆θP2 = −6 (deg).) Calculated phasebias values are shown in each graph.

55

150

Time (ps)150

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Allocate Nbuf -word memories for(segmented) time profiles ofcarrier density nseg(i) andcontrol pulse intensity Pseg(i)loop index iloop=0
iloop = Nloop?

load SOA and othersimulation parameters(total bit length Nbit,
bit-rate B, time step δT,
DGD ∆TDGD, T1st-bit,)

Nstep=Nbuf-1

Nstep=Nres-1load P(k) ~ P(k+Nstep) to Pseg(i)nseg(0)=ninitload bitpattern C(l), l=0~Nbit-1

index i=0 to Nstep-1
determine loop number & residual steps,
calculate G(i), ∆Φ(i),t(i)Eout(i),Pout(i)Ntotal = (Nbit/B+T1st-bit)/δT
Ntotal = Nbuf*Nloop+Nres

save results to the hard disk
set P(i) to be 0
calculate initial carrier density ninit
Allocate nprev(j) (buffer for previous
loop results) and set to ninit

calculate δn(i)using 2nd-order
Runge-Kutta methodnseg(i+1)=nseg(i)+δn(i)create long P(i) profile, i=0~Ntotal-1, using C(l)i=i+1
ninit=nseg(Nstep)

save P(i) to hard disk

store nseg to nprev(j)iloop=iloop+1
Free buffers andend simulation
Figure 3.9: Flow chart of our DISC-gate waveform simulation.

56

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

50

150

Time (ps)

(c)
1000
300
100
30
10
3
1
0.3
0.10

100

Signal intensity (arb.unit)

1000
300
100
30
10
3
1
0.3
0.10

(b)

∆ΦB =1.075π∆ΦB =1.060π

50

100

150

Time (ps)Signal intensity (arb.unit)

Signal intensity (arb.unit)

Signal intensity (arb.unit)

(a)

1000
300
100
30
10
3
1
0.3
0.10

(d)
1000
300
100
30
10
3
1
0.3
0.10

∆ΦB =1.030π

50

100

Time (ps)∆ΦB =1.100π

50

100

Time (ps)Figure 3.10: Calculated cross-correlation traces of the DISC output: (a) with 25GHz input pulses, (b)-(d) with 12.5-GHz pulses. ((b) ∆ΦB = 1.030π, (c) ∆ΦB =1.060π, (d) ∆ΦB = 1.100π). Other parameter values are summarized in Tables 3.2
and 3.3.

57

150

150

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Parameterf
TFWHM

EpulsePcw∆t
Tprobeνcw
νpulseTable 3.2: Parameters used in the DISC-gate simulation
Description
Value
Value
Pulse repetition frequency
Input pulse width (FWHM)
Input pulse energy (chip)
Input cw power (chip)
Delay time in the MZICorrelator probe width
Frequency of cw light
Frequency of pulsed light

Value(Fig. 3.10(a))
25 GHz(Figs. 3.10(b)-(d))
12.5 GHz(§3.4.2)
10.5 GHz (50 GHz)

3.8 ps429 fJ
5 mW
5.0 ps2.2 ps
193.7 THz
192.1 THz

3.8 ps1129 fJ
5 mW
5.0 ps2.2 ps
193.7 THz
192.1 THz

2.0 ps
30 fJ
0.3 mW
5.0 ps2.0 ps
193.7 THz
192.8 THzshown in Figs. 3.10(a) to 3.10(d), after choosing ∆ΦB values properly. The calculated relative peak intensities of the sub-pulses were −24 dB in Fig. 3.10(a) and−18 dB in Fig. 3.10(b).
For more systematic comparison of the measured and calculated results, though,we studied how the relative intensities of sub-pulse-like components behave as functions of the MZI phase bias ∆ΦB (Fig. 3.11). The relative intensity was taken at
three time positions (t = t0 + 15 ps, t0 + 40 ps, and t0 + 65 ps, where t0 is the primary pulse’s position). The calculated behavior was quite similar to the measured
results, taking the ∼ 27-dB dynamic range of the cross-correlator into account.
The mismatch in the intensity was small, 1 ∼ 2 dB for t = t0 + 15 ps series, at∆ΦB = 1.00π ∼ 1.08π. We are aware that the phase-bias dependence and extinction ratios are dominated by many parameters, and future improvement in the gatemodel may solve the remaining small mismatch. Regarding sub-pulse intensity, we
consider that the gate model in this chapter provides the most reliable predictions
currently available.
Thus, as shown in Figs. 3.8-3.11, we observed a good match between the measured and calculated waveforms and the phase-bias dependence. Therefore, we presume that our sub-pulse generation model has been experimentally veriﬁed for theparticular recovery rate, and that we can predict sub-pulse intensity with 1 ∼ 2 dB
accuracy using this model.

58

(a)
0
-10
-20
-30
-40
1.00

1.04

1.08

∆ΦB /π1.12

Relative intensity (dB)

Relative intensity (dB)

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

(b)
0
-10
-20
-30
-40
1.00

1.04

1.08

1.12

∆ΦB /πFigure 3.11: Relative intensities of sub-pulse-like components in 12.5-GHz DISC
output: (a) from experimental results, (b) from calculation. These were obtainedat three timing positions: t = t0 + 15 ps (solid), t0 + 40 ps (dashed), and t0 + 65 ps(dotted).

3.4.2

Trade-oﬀ regarding sub-pulse generation and patterninduced intensity ﬂuctuation

The sub-pulse intensity of the DISC output discussed in §3.4.1 could be suppressedby tuning the MZI bias down to ∼ −20 dB, which was negligible for practicalapplications. Nevertheless, we have regarded sub-pulse generation as a major issue
in the case of random-data operation, because of the trade-oﬀ relationship between
sub-pulse generation and PIF. If the input signal is not a continuous clock but has arandom pattern, the instantaneous carrier density in the SOA varies upon each signal
injection depending on the preceding input-signal pattern. Then the intensitiesof the output signals become pattern-dependent. This eﬀect can be reduced byaccelerating the carrier recovery with current injection or assist light injection. Our
sub-pulse model indicates, however, that a sub-pulse becomes large as we increase
the carrier recovery rate. In this section, we will discuss the measured and calculated
results for this eﬀect.
We systematically measured sub-pulse waveforms for several carrier lifetimes,
using the customized SOA chip B#1, as explained in §3.3.1. Figure 3.12(a) shows
the gain-recovery proﬁles of the SOA for several injection currents, which we showed
in Fig. 2.9(c) in Chapter 2. The recovery time τC was reduced from 280 to 36 ps as we
increased the injection current Iop from 100 to 400 mA, and it was approximately
inversely proportional to Iop above 150 mA. The eﬀective recovery time τeﬀ was
further shortened to 11 ps after we injected 80 mW assist light with a nearly59

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER(b)
400 mA + assist light (τeff= 11 ps)

0

IOP = 400 mA (τc= 36 ps)
300 mA (τc= 40 ps)-5200 mA (τc= 75 ps)
150 mA (τc= 117 ps)
100 mA (τc= 281 ps)

-10
-200

0

+200

Time (ps)+400

Signal intensity (arb.unit)ln (XGM output) (arb.unit)(a)10
400 mA with assist

1
400 mA
200 mA

0.1

150 mA

0.01
0.001

100 mA

0

+50

+100

Time (ps)Figure 3.12: DISC output waveforms for several carrier recovery rates. (a): XGM
proﬁles of the SOA B#1 for several operation currents, indicating the recovery
acceleration. The cw-probe intensity and pulse energy into the chip were set to−25 dBm and 10 fJ, respectively. Note that each plot is translated intentionally invertical direction. (b): 10.5-GHz DISC output waveforms. ∆ΦB was set to be 1.0π.
transparent wavelength (1480 nm) into the chip. Using this SOA, we measured the
DISC output waveforms on each Iop . The operation frequency was 10.5 GHz. The
input pulse width, wavelength, and energy into the chip were 2 ps, 1555 nm, and
30 fJ, respectively. The cw light wavelength and power into the chip were 1548.5
nm and −5 dBm, respectively. The delay time of the MZI was set to 5 ps. The
MZI phase bias was ﬁxed at 1.0π throughout the measurement, since the optimum
phase bias depends largely on τC (τeﬀ ). The measured waveforms are shown in Fig.
3.12(b). We found that the relative intensity of the sub-pulse increased up to −10
dB, as we increased the carrier recovery rate 1/τC (Fig. 3.13(a)).
Meanwhile, we could predict this result qualitatively using our gate model. For
simplicity, we assumed that τC was inversely proportional to Iop and other fundamental parameters did not change with Iop . The adopted values are summarized inTables 3.2 and 3.3. The values in Table 3.3 were determined in view of measured
characteristics of SOA B#1 at Iop = 400 mA. Then the measured and calculated
sub-pulse intensities agreed quite well for most 1/τC values (Fig. 3.13(a)). We regarded this matching as convincing evidence that our model is applicable in a wide
1/τC range. Some discrepancy at τC = 11 ps probably originated from the diﬀerence
between the acceleration mechanism with current injection and that with assist-light
injection. This result clearly indicates that we can suppress sub-pulse generation aslong as we keep the carrier recovery rate small.
As we mentioned above, this method of sub-pulse suppression leads to an increase60

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

Table 3.3: SOA fundamental parameter values used in the calculation, and derived
quantities
Parameter
Description
Value in §3.4.1
Value in §3.4.2
Iop
Injection current
250 mA
400 mA ×α
Leﬀ
Active region length (eﬀective)
700 µm1100 µm−103VActive region volume
1.4 × 10(cm )5.2 × 10−11 (cm3 )Γ
Conﬁnement factor
0.2
0.2
τC
SOA carrier lifetime
200 ps35 ps /αdg
SOA diﬀerential gain
2.17 × 10−15 (cm2 )6.36 × 10−16 (cm2 )dn
ηTCarrier-conversion eﬃciency
0.079
0.34
dnr
dn
nmax
c

G0
sat
Epulse∆Φmax

Refractive index change
Maximum-available carrier density
SOA small-signal gain
Saturation energy for pulse
Phase shift at complete carrier depletion−1.87 × 10−19 (cm3 )1.76 × 1017 (cm−3 )−1.4 × 10−20 (cm3 )5.72 × 1017 (cm−3 )23.4 dB
600 fJ+6.0π

34.8 dB
473 fJ+2.3π

in PIF. We calculated the DISC output waveforms for a 50-Gb/s input pulse train
with a pseudorandom bit sequence (pattern length = 215 − 1), and ﬁgured the PIF
from the intensity ratio of the largest and smallest signals in the output pulse train(Fig. 3.13(b)). The other parameter values were kept unchanged from the former
calculations. The dotted line in Fig. 3.13(a) shows the dependence of PIF on therecovery rate. Evidently PIF could be suppressed by increasing recovery rate, whichlead to the unwanted increase in sub-pulse intensity.
In this way, our model shows trade-oﬀ between sub-pulse generation and PIF.
Although what we have shown is only a particular example, we consider that thistrade-oﬀ relation arises fairly universally. We will encounter sub-pulses as large as−7 dB, in the present case for example, as we suppress the PIF down to 1 dB(or −10 ∼ −12 dB when PIF is 4 dB). The situation may become even worse
depending on other operating conditions such as signal bit rate, signal duty ratio,
and the control pulse energy. Therefore, we consider that a new gating scheme tocoping with this issue is necessary, and that sub-pulses should be evaluated using
the current model, or an expanded model in the near future.
Here we mention a few things. In some reports on the results of the ultrafast
wavelength-conversion experiments[23, 26], we cannot ﬁnd such large sub-pulses. We
suppose several possible reasons for this.
1. PIF was not highly suppressed, as indicated by the eye-diagrams (2 ∼ 3 dB?).
61

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

+15
+12

-10

+9
-15
+6
-20
-25

1

+3

5 10
50 100Carrier recovery rate (GHz)0

PIF (dB)

Sub-pulse intensity (dB)-5Signal intensity (arb. unit)

(b)

(a)

40

τc = 14 ps

30
20
10
0
8
6
4
2
0

τc = 70 ps

-20

0
+20
Time (ps)+40

Figure 3.13: Trade-oﬀ between the pattern-induced intensity ﬂuctuation (PIF) andthe sub-pulse noise. (a): Measured (solid) and calculated (dashed) sub-pulse intensities against the carrier recovery rate 1/τC at 10.5 GHz, and calculated PIF for the
50 Gb/s signal (dotted). (b): Calculated eye diagrams of the DISC output for 50
Gb/s pseudo-random signal, for two carrier recovery rates. Cross-correlation process
was omitted for simplicity. Parameter values are summarized in Tables 3.2 and 3.3.
2. The MZI phase bias values were detuned from π, and the sub-pulses were
suppressed to some extent (Fig. 3.12).
3. Unexpected contribution of the nonlinear polarization rotation (as in Chapter
4 or as in [57]).
4. Unexpected contribution of the carrier heating [62] and the BPF detuning(Chapter 4).
Since there were many potential factors in the past experiments, it is diﬃcult toprovide clear answer to them. In Chapter 4, however, we still observe unexpectedlysmall sub-pulses when no BPF-detuning is employed. We are not sure this is accidental or intrinsic. We speculate that the contribution of the carrier heating is quite
large in the frequency region over 160 GHz, and some modiﬁcation in our model will
reduce the mismatch in such high-frequency region.

3.5

Conclusion

We have studied sub-pulse generation of the ﬁrst kind (originated from the exponential recovery proﬁle of the SOA carriers) in the wavelength-converted signal
62

CHAPTER 3. OBSERVATION OF SUB-PULSE GENERATION FROM THE
DISC-TYPE WAVELENGTH CONVERTER

from a DISC gate operated at 10 ∼ 25 GHz, both theoretically and experimentally.
We observed that the DISC output contained small sub-pulses in addition to thewavelength-converted output pulses. Good matching (1 ∼ 2 dB) was obtained between the measured and calculated sub-pulse intensities. The measured dependenceof sub-pulse waveforms on the MZI setting was quantitatively explained with our
DISC model.
In addition, we demonstrated that our model is applicable in a wide range of SOA
carrier recovery rates. The impact of carrier-recovery acceleration (from τC = 280
ps to τeﬀ = 11 ps) on sub-pulse waveforms was systematically studied using a DISC
with a customized SOA chip. We obtained good agreement between the measured
and calculated sub-pulse intensities for the majority of the recovery rate range. We
consider that our model is quite reliable as long as the inﬂuence of the carrier-heatingin the SOA is small.
Using our sub-pulse model, we discussed the trade-oﬀ between sub-pulse generation and PIF. Our calculations indicated that signiﬁcant sub-pulses may emerge
when we suppress PIF, which may lead to diﬃculty in the practical high-speed operation of the DISC. This result suggests that part of the DISC structure should be
improved to overcome this trade-oﬀ. We are in the process of solving this issue using new gating schemes, for example, using an expanded DISC including an optical
spectral synthesizer[62].
We believe that the success in such modeling will be valuable not only for thepractical applications of the DISC but also for the detailed understanding of otherkinds of all-optical gates using SOAs, delayed interferometers, and so on.

63CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

Chapter 4200-Gb/s wavelength conversion using DISC withlong-pattern waveform monitoring

4.1

Introduction

Through chapter 2 and 3, we have investigated the characteristics of SOA-based alloptical gates and DISCs. We have tried to understand their characteristics in thefrequency range above 100 Gb/s, as well as in the lower frequency range as ∼ 10 GHz.
Since we had to analyze the properties of the SOAs and the DISCs as accurate as
possible, we used the experimental results precisely measured at the lower frequency
range. For the comprehensive understanding of the gate characteristics, however,systematic study on the high-frequency operation of all-optical gates will be also
desired.
In this chapter, we discuss our results on the output waveform measurements ofa DISC gate at 200 Gb/s. As we have shown through Chapter 3, the waveform monitoring is a powerful method to understand the fundamental dynamics of all-opticalgates. So far error-free operations of DISCs at 160 Gb/s have been reported by a
few groups[23–26], and waveform monitoring has been performed in several ways.
Each of the methods had their own drawbacks, however. The most common method
was to use a set of optical time-division demultiplexing gates (DEMUXs), OE converters and electric sampling scopes. In such measurements, detailed information of
the waveforms was lost due to the bandwidth limits of the electric devices. Costly
streak-cameras were sometimes used, but their time resolutions were also limited
to ∼ 4 ps. Eye-diagram monitoring of the output waveform has been reported forthe extended DISC operated at 160 Gb/s and 320 Gb/s using an optical sampling
scope. Though it has an excellent time resolution[78], it is based on (what is called64

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING) “asynchronous sampling” [79, 80] and not suited for the characterization of eachpulses in a patterned data signals. Result on the systematic measurement has notbeen reported from them, neither. Finally, we measured and reported the inﬂuenceof ultra-fast eﬀects on the patterning eﬀect of an SOA gate (similar to DISC) at
105 Gb/s, through the cross-correlation method[43]. The repetition-rate of the bitpattern allowed in this method was restricted by the frequency of the probe pulse(10.5 GHz), however, and the adopted bit-pattern (‘11110000001111000000· · ·’) was
far from realistic digital data. We performed a 1/16-demultiplexing of the probe
pulses using an LN modulator and a programmable pattern generator (PPG) in
Chapter 2. This method requires, however, an exclusive use of the costly PPG. The
extinction ratio of the demultiplexed pulse was not quite high, either. Therefore wedeveloped a low-frequency (40 MHz) sampling-pulse generator with low additional
cost and used it as a probe source to the cross correlator. This enabled us to acquire
the waveform of very-long (Tperiod = 25000 ps, 4992 bits at 200 Gb/s) bit-pattern
digital signal. Through this long-pattern waveform measurement, we aimed to investigate the pattern-induced intensity ﬂuctuation (PIF) of the output signal andother waveform distortions, more strictly than in the preceding short-bit[42, 43] or
27 − 1-bit[23, 25, 26] experiments.

4.2

Experimental setup for the 200-Gb/s wavelength conversion

Figure 4.1(a) shows the overview of our experimental setup. In the following sections, we ﬁrst discuss the data-multiplexing part (Fig. 4.1(b)), the sampling-pulse
generator (Fig. 4.1(d)), and the measured waveform of 200-Gb/s input-data signal.
Then we provide measured results on the 200-Gb/s operation of the DISC gate (Fig.
4.1(c)).

65

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

bit-pattern signal

(a)
input

SOA gate
outputSynch. RF

12.48-GHz
synthesizer

200 Gb/s
multiplexing

MLFL #1

PPG

BPF

9 nm

EDFA

cross
correlator

sampling pulse
40 MHzgenerator
pulses, 600 fs(b)

bit-pattern signal
Power
PC meter

1.6 psP EDFAPC

DDF #1VDL

EDFA
DT2=6.25 ns

DT1=12.5 ns

VDL

VDL

OTDM
MUX
DCF

P
DT3=6.4 ns

monitor PD

LN modulator #1

VDL

P

RF amp #1

DT4=3.2 ns

MLFL #1
MUX 3&4 MUX 1&2
MLFL #2

PPG
SMF 100 m

66

DDF #2
DDF #1EDFA

VOACHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(c)

Power
meter

λ2=1540 nm
PC
DFBλ1=1558 nm LD
200 Gb/s
EDFA PC

DISC-Gate

Power
meter

MZI

SOA

BPF
9 nm

λ2=1540 nm
OutputQ1 H1 P1 C1 Q2 P2200 Gb/s
BPF∆T= 1 ps
9 nm

Power
meter

(d)
12.5 GHz
pulses, 1.6 ps

LN modulator #2

VDL

EAM

RF Amp #2

BPF
BPF
DDF #2 9 nm
or 7.5 nm
3 nm

40 MHzpulses,
1.6 psRF Amp #3

EDFA

Photo
electric pulse,
diode
40 MHz, ~30 ps
16 GHzmonitor PD
VOA

Sync. RF12.5 GHz

Freq. divider
N = 312Mode-locked
fiber laser #2
Sync. RF
40 MHzEDFA

40 MHzSMF
300 fs,
100 m
Jitter~ a few ps

67

VDL

40 MHzpulses, 600 fs
low jitter,high extinction

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING(e)

Figure 4.1: The setup of the 200-Gb/s DISC-gate experiment. (a): Experimental
setup overview. (b): Diagram and photograph of the optical time-division multiplexing part (c): DISC gate (d): Simpliﬁed diagram of the 40-MHz sampling-pulse
generator (e) Photograph of the core part of the sampling-pulse generator

68

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

4.2.1

Optical time-division multiplexing of mode-locked pulses
up to 200 GHz for the gate input

The time-division multiplexing of RZ-formatted optical signals is possible by splitting them into two components, applying proper temporal shift to one component,
and combining them after that. We multiplexed the repetition rate of mode-locked
pulse trains (λ = 1555 nm) from 12.48 GHz up to 12.48 × 24 = 199.68 GHz, using
four-stage multiplexers (MUXs). The ﬁrst and second MUXs have been fabricatedby ourselves using commercially-available ﬁber-optic devices, while the latter two
were prefabricated MUX products (Pritel Inc., OCM-4). All the devices in our
MUXs were pig-tailed by PM-DSFs. Since the third and fourth MUX had certain
amount of velocity dispersion, though, we had to use a dispersion-compensation
ﬁber (β2 = −84 ps/nm·km, β3 = −0.26 ps /nm2 ·km) of 1 m after multiplexing.
Since our target frequency was as high as 200 GHz, we performed adiabatic
compression[81] of the 12.48-GHz pulses before multiplexing, using a dispersiondecreasing ﬁber (DDF) module. Figures 4.2(a) and 4.2(c) show auto-correlation
traces of the pulses before and after compression in the best case. We see that thepulse width was shortened from 1.6 ps to 260 fs, when the average input power
was +21.3 dBm. The waveform of the compressed pulse generally depends on thestatus of the input mode-locked pulses, and hence we had to tune the operating
condition of the MLFL carefully from day to day. Note that the correlation trace ofthe neighboring pulses in the 12.48-GHz pulse train, as shown in Fig. 4.2(d), wasslightly broadened from that of the identical pulses, probably due to an instabilityof our MLFL. This broadening was one cause of the bit-rate limitation in our crosscorrelation measurements.
Figures 4.3(a) to 4.3(d) show auto-correlation traces of the multiplexed pulse
trains. Periodic traces were obtained after ﬁne adjustment of the delay length ineach MUX stage. In general, we had to perform this adjustment on each day, at least
for the 1st and 2nd MUX stages. The group velocity dispersion, which broadened
the pulse width at 100 GHz to 1.36 ps, was partially compensated by the DCF at
200 GHz. The resultant pulse-width of 800 fs was suitable for 200-Gb/s operationof the DISC gate, in view of the ﬁnite time resolution of our sampling system.
Figure 4.4 shows the cross-correlation trace of the 200-GHz clock pulses and 12.48GHz compressed clock pulses. The intensities of the pulses in the sixteen OTDM
channels were adjusted with the variable attenuators, so as to be almost equal witheach other.
69

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORINGSHG Intensity (arb. unit)

2.0
1.5

(a) Before compression (b)

1.0
0.5
0.0
2.0
1.5

(d)
(c) After compression

1.0
0.5
0.0

-4

-2

0

+2

+4 -84

Time (ps)Intensity (dBm/nm)

-80

-78

-76

Time (ps)
(f)

(e)
+20

-82

1558 nm

1555 nm

0-20-40Before compression
-60
1545

1550

1555

1560

After compression

1500 1520 1540 1560 1580 1600

Wavelength (nm)Wavelength (nm)Figure 4.2: Auto-correlation traces and optical spectra of the 12.48-GHz modelocked pulses before and after compression. (a), (b): Before compression, aroundt = 0 ps and t = −80 ps. (c), (d): After compression, around t = 0 ps and t = −80
ps. (e) and (f): Typical spectra of the pulses before and after compression.

70

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORINGSHG Intensity (arb. unit)

1.5

(a)

25 GHz, width ~220 fs(b)

50 GHz, width ~260 fs(c)

100 GHz, width ~1.36 ps(d)

200 GHz, width ~800 fs

1.0

0.5

0
0.3

0.2

0.1

0
-150

-100

-50

0

+50

-100

Time (ps)

-50

0

+50

+100

Time (ps)Figure 4.3: Auto-correlation traces of the multiplexed pulse trains. (a): at 25 GHz.
(b): at 50 GHz. (c): at 100 GHz. (d): at 200 GHz.
Bit-pattern encoding was performed at 12.48 GHz through an LN modulator(#1). Since the frequency of our new sampling-pulse generator was restricted to
40 MHz, we chose a bit-pattern with a pattern length of 312 as shown in Fig. 4.5
and synchronized this pattern to the sampling pulses. The bit-pattern in this stage
was simply determined through a Monte Carlo method. To avoid the temporalcorrelations of multiplexed bits in a short span, we inserted an additional ∼ 12.5-ns
delay in the ﬁrst MUX and a ∼ 6.25-ns delay in the second stage. The delay values
in the third and fourth stages were pre-adjusted for 10-Gb/s, 27 − 1-bit PRBS input
signal and could not be changed.

71

SFG intensity (arb. unit)

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

0.5
0.4
0.3
0.2
0.1
0

-60

-40

-20

0

Time (ps)Figure 4.4: Cross-correlation trace of the 200-GHz pulse train with 12.48-GHz compressed probe pulses.

1

0
0

24

48

72

96

120 144 168 192 216 240 264 288 312

Bit index
Figure 4.5: Adopted encoding pattern at 12.48 Gb/s.

72

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

4.2.2

Generation of low-jitter sampling pulses at 40 MHz

A low frequency, low-jitter, distortion-free, ultrafast and high-power pulse source
is inevitable for the optical sampling of long-period, high-bit-rate lightwave signals. Several commercially-available MLFLs satisfy some of these requirements. In
fact, we have a custom-designed MLFL (Calmar Optocom Inc., FPL-02CFTUEC12)
which generates low-distortion, ultrafast (FWHM = 200 fs) optical pulses at 40-MHz
repetition frequency (Fig. 4.6(a)). It is not easy in general, however, to synchronize
such low-frequency pulses to other ultrafast signals with small timing jitter. Figure
4.6(b) shows a measured example of several cross-correlation traces of compressed
pulses from the 12.48-GHz MLFL (MLFL #1) and pulses from the 40-MHz MLFL
(MLFL #2), acquired in a short period. The experimental setup was as shown in
Fig. 4.6(c). The correlation traces show that the timing jitter between pulses fromtwo MLFLs exceeds a few picoseconds. Furthermore, temporal position of the pulses
from MLFL #2 drifted in a few hours by several hundred picoseconds, presumably
due to the thermal expansion of the laser cavity. Thus we had to develop anotherlow-frequency pulse generator which can be easily synchronized to the 12-GHz system.
The setup of our sampling-pulse generator was as shown in Fig. 4.1(d). Thissystem requires a branch of the 12.48-GHz pulse train and a synchronized electric
signal for the input. The pulse train was electrically demultiplexed down to 40
MHz, through a 10-Gb/s LN modulator (#2) and a 40-Gb/s EA modulator (OkiElectric Industry, OM5642W-30B). To generate high-speed electric gate signals forthese modulators, we used the 40-MHz MLFL and a 16-GHz photo-diode (Discovery
Semiconductor, Inc., DSC20H). Figure 4.7(a) shows the OE-converted signal of themode-locked pulse, acquired with a 20-GHz electric sampling scope. The averaged
input power to the photo-diode was typically adjusted to −14 dBm, though Fig.
4.7(a) was exceptionally acquired at −20 dBm. This gate signal was split into two
by an Wilkinson divider, and ampliﬁed by broadband (∼ 12 GHz) RF ampliﬁers asshown in Figs 4.7(b) and 4.7(c). The two ampliﬁers had fairly large diﬀerence intheir performances. The ﬁrst inverting ampliﬁer #2 (Cernex Inc., CBMV3142019)
seemed to have poor broadband-gain-ﬂatness, and the generation of the after-pulse
could not be avoided. Since the EA modulator did not accept inverted gate pulses,
this ampliﬁer was used for the LN modulator. The output signal from the second
ampliﬁer (Picosecond Pulse Labs. Inc., 5865) was less distorted, and used for the
EA modulator. Figures 4.7(d) and 4.7(e) show the gate windows of the LN and EA

73

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

modulator driven by these electric pulses, measured using cw-light input, a 35-GHz
photo-diode, and the sampling scope. The width of the windows, broadened by the
band-width limits of the photo-diode, ampliﬁers, and modulators, were much larger
than the timing jitter of the MLFL #2, and still smaller than the interval of the
12.48-GHz pulse train. The FWHM length of the cascaded windows measured using
the auto-correlator was about 30 ps. As the 12.48-GHz pulse train went throughthese two modulators, the pulses outside the cascaded windows got suppressed withan estimated extinction ratio of ∼ 30 dB. The modulated waveform after LN #2
was monitored by the sampling scope during the experiment, so that easy tuning of
the DC bias for LN #2 and the timing of the gate windows was possible.
The pulses right after the demultiplexing were fairly weakened due to the insertion losses of the EA modulator and other devices. Therefore we ampliﬁed them
twice with dispersion-compensated EDFAs. After then we performed adiabatic compression using another DDF module (#2), and ampliﬁed the compressed pulses
again. Figure 4.8(a) and 4.8(b) show the auto-correlation traces of the sampling
pulses right after the third EDFA. The pulse width was compressed down to 440 fs.
This is almost the shortest value we could achieve. The suppression ratio of the peakat t = −80 ps was 19 dB. This probably resulted from the after-pulses in the gatewindows, and we expect that most of the pulses in the 12.48-GHz pulse train were
suppressed with higher ratios. Figure 4.8(c) shows the spectrum of the generator
output after the third EDFA. The broad peak around λ = 1540 ∼ 1570 nm indicate
the spectrum of the compressed 40-MHz pulses, which was broadened by self-phasemodulation inside the DDF, while the un-broadened peak at λ = 1555 nm partiallyresulted from residual 12.48-GHz pulses. To reduce the contribution from the residual 12.48-GHz component, we used a 9.2-nm (or 7.5-nm in §4.3) bandpass ﬁlter. The
autocorrelation traces and optical spectra in these cases were as shown in Fig. 4.8(d)to 4.8(f). Though the pulse width was broadened to 620 fs (BPF 9.2 nm) or 740 fs(BPF 7.5 nm), the extinction ratio was improved to at least 27 dB. This extinction
ratio is comparable or larger than the demultiplexing ratio (12480/40 = 312), and isenough for the cross-correlation measurements of patterned signals in the following
sections.

74

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORINGSHG Intensity (arb. unit)(a)1

pulse width
200 fs

0.8
0.6
0.4
0.2
0

-4

-2

0

+2

+4

Time (ps)(c) 12.48-GHz 12.48 GHz(b)synthesizer

SFG Signal (arb. unit)

0.2

MLFL #1

DDF #1

EDFA

0.1

Freq. divider
N = 312

MLFL#2

Sync. RF
40 MHz0+20

600 fs

2 ps

+40

cross
correlator

40 MHzpulses, 200 fs+60

Time (ps)Figure 4.6: The auto- and cross-correlation traces of the mode-locked pulses from
MLFL #2 and a schematic diagram of this measurement. (a): Auto-correlation
trace. (b): Several cross-correlation traces of pulses from MLFL #1 and #2, taken
within a few seconds. (c): Schematic diagram of the cross-correlation measurementin (b).

75

Voltage (V)CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

+0.4
+0.2

Voltage (V)PD output(b)

RF amp #2

0
-0.2
-0.4
+0.4
+0.2
0
-0.2
-0.4
+0.4

Voltage (V)(a)(attenuated by 20 dB)(c)

RF amp #3

+0.2
0
-0.2

(attenuated by 20 dB)

Transmittance(arb. unit)

+1.0
+0.8
+0.6
+0.4
+0.2

Transmittance(arb. unit)

-0.4

+1.0
+0.8
+0.6
+0.4
+0.2

(d)

LN window

0(e)

EAM window

0

-200

-100

0

+100 +200 +300

Time (ps)Figure 4.7: Electric gate-signal generation in our sampling system. (a): OEconverted signal of the pulse from MLFL #2. (b),(c): Signals ampliﬁed by theRF ampliﬁers. (d),(e): Gate windows generated with the LN and EA modulators.
Superposed gray curves indicate the 12.48-GHz pulse train to be modulated.
76

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(d)

(a)

SHG Intensity(arb. unit)

SHG Intensity(arb. unit)

3

2

1

0
-4

-2

0

+2

0.8
0.6
0.4
0.2
0

+4

-4

Time (ps)

SHG Intensity(arb. unit)

SHG Intensity(arb. unit)

0.3
0.1
0.03

+4

1
0.3
0.1
0.03

0.01

0.003
0.001

0.01
0.003
0.001
-50

0

+50

0.00030.0001-100+100

Time (ps)

(f)

-50

0

+50

+100

Time (ps)+20

Intensity (dBm/nm)+20

Intensity (dBm/nm)

+2

(e)

3
1

(c)

0

Time (ps)

(b)

-100

-2

0
-20
-40
-60

0
-20
-40
-60

-80
1480 1500 1520 1540 1560 1580 1600 1620

-801480 1500 1520 1540 1560 1580 1600 1620

Wavelength (nm)Wavelength (nm)Figure 4.8: Auto-correlation traces and optical spectra of the ultrafast pulses obtained from the sampling pulse generator. (a) to (c): without BPF after the last
EDFA, (d) to (f): with 9.2-nm BPF (solid) and 7.5-nm BPF (dashed). (a),(d): Shortspan, linear scale. (b),(e): Long span, logarithmic scale. (c),(f): Optical spectrum
measured before the last band-pass ﬁlter. Resolution was 0.5nm.

77CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

4.2.3

Waveform monitoring of long-pattern 200-Gb/s optical signal

Before we perform 200-Gb/s operation of the DISC, we tested our sampling systemwith the 200-Gb/s input signal.
Figure 4.9 shows the cross-correlation trace of the 200-GHz clock pulses with
40-MHz sampling pulses. From the width of each pulse in the trace (1.5 ps), we
estimated the time resolution of our measurement system to be slightly better than
1.5 ps. Constant background degraded the extinction ratio to ∼ 6.5 dB (from 12.2
dB in the case of Fig. 4.4), but still we could observe a clear waveform.
Then we measured the patterned waveform of the 200-Gb/s signal, encoded at
12.48 GHz. Since the time span of our cross-correlator was limited to ∼ 150 ps,
we could measure only a short division of the 25000-ps-long pattern waveform ina single scan. By applying a bit shift to the encoding bit-pattern at 12.48 GHz
for every single scan, we successfully monitored the long-pattern waveform over thewhole period. The entire waveform will be found in the appendix. Figures 4.10(a) to4.10(g) show typical examples of the patterned waveform in the long-pattern signal.
In some sections, ‘1’ bits and ‘0’ bits appeared alternately. In some other sections,
successive ‘0’ or ‘1’ appeared. Maximum length of the successive ‘0’ in the whole
period was eleven, and that for ‘1’ was eight. Figure 4.11 shows the (temporal)correlation of the bit-pattern, deﬁned as∑ {nperiodC(∆k) ≡}
(B(k) − B) × (B(k + ∆k) − B)

k=0

∑

(4.1)

nperiod(B(k) − B)2k=0

where B(k) represents the on-oﬀ of the k-th bit and B represents the average of B.
This result indicates that the bit-pattern we used was close to random, though notcompletely random as in the case of the widely-used ‘pseudo-random’ bit pattern.
Note that we paid high attention to less-random parts (e.g. ‘00000000001’) in thewhole bit-sequence, rather than the randomized parts, in order to evaluate PIF.
Figure 4.10(h) shows an eye-diagram-like expression of the input signal, obtainedusing 158 piece of the patterned correlation traces ∗) . Careful compensation of the∗)

Not the actual eye-diagram, since each dot in this diagram was obtained after averaging many
data.

78

SFG intensity (arb. unit)

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

0.02

0.01

0.00

-60

-40

-20

0

Time (ps)Figure 4.9: Cross-correlation trace of the 200-GHz clock pulses with 40-MHz sampling pulses. The gray line shows a background output, which was acquired byinjecting only sampling pulses into the correlator.
timing drift between the traces has been achieved through data analysis, and we
successfully obtained clear eye-opening.

79CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

0.04
0.03

(a)

(b)

(c)

(d)

(e)

(f)

SFG Intensity (arb. unit)

0.02
0.01
0.00
0.03
0.02
0.01
0.00
0.03
0.02
0.01
0.00

(h)

(g)
0.03

0.04

0.02

0.03

0.01

0.02

0.00 0 +20 +40 +60 +80 +100 0.01

Time (ps)

0
-60

-40

-20

0

+20

Time (ps)Figure 4.10: Cross-correlation traces of the 200-GHz patterned signal with 40-MHz
sampling pulses. (a)∼(g): Examples of the traces measured in single scans, withcertain bit-shifts. (h): Eye diagram of the 200 Gb/s input signal, calculated using
the measured correlation traces.
80CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

Figure 4.11: Correlation of the multiplexed bit-pattern.

4.3

Wavelength conversion with DISC at 200 Gb/s

The DISC gate we investigated was almost same as that we used in §3.4.2. In viewof the short pulse-width required for the 200-Gb/s operation, we replaced the 3nm bandpass ﬁlter after the SOA to that with 9.2-nm width, and the calcite with
5-ps DGD to that with 1-ps DGD. The wavelength of the cw light was changed
from 1548.5 nm to 1540 nm because of the larger signal bandwidth. We spared theB#1 SOA sample for this experiment and used another sample with same structure
instead (B#4). A half-wave plate was necessary before the MZI (ﬁg. 4.1(c)), since
our calcite module in this experiment was not rotatable as in the case of Chapter3.
The MZI bias was set around π.
We operated the DISC gate with the 200-Gb/s signal described in the previous
section, and monitored its output signal with the 40-MHz sampling system. Figures4.12(a) and 4.12(c) show typical input and output waveforms. We see that the
on-oﬀ patterns of those two were same, while the output waveform shows large
PIF. Figures 4.12(b) and 4.12(d) show the input and output optical spectra. The
center wavelength of the input signal was shifted to 1558 nm from that of the initialmode-locked pulses (1555 nm), through self phase modulation in the DDF. The
center wavelength of the output was 1540 nm, determined by the cw light fromthe DFB-LD and the bandpass ﬁlter detuning. While the input signal spectrum

81CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(b)

0.04

Intensity (dBm/nm)

SFG Intensity (arb. unit)

(a)

0.03
0.02
0.01
0
-100

0

+100

0

-20
-30
-40
-50
-60
1550

+200

Time (ps)

1560

1565

1570

(d)

0.15

0

Intensity (dBm/nm)

SFG Intensity (arb. unit)

1555

Wavelength (nm)

(c)

0.1

0.05

0
-100

res = 0.05 nm

-10

0

+100

+200

Time (ps)res = 0.02 nm

-10
-20
-30
-40
-50
-60
1530

1535

1540

1545

1550

Wavelength (nm)Figure 4.12: Typical input and output signals of the DISC. (a) Part of the input
waveform (60 bits out of 4992 bits). (b): Spectrum of the input signal. (c) Corresponding part of the output waveform. The SOA injection current was 200 mA, thecw power into the chip was −5.3 dBm, and the pulse energy into the chip was 3.3
fJ. (d) Spectrum of the output signal.

82

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

looks continuous due to unstable characteristics of the MLFL, the output spectrum
clearly shows discrete components separated by 200 GHz. 2.2-THz up-conversion of
the carrier frequency was achieved.
Figure 4.12(c) indicates that the primary issue regarding the DISC operationis the PIF. Therefore we will focus on it from now. Our deﬁnition of PIF in thissection is as follows: First, we assumed that the strongest output pulse appears
around the bit sequence of ‘0000100000000001’. (Fig. 4.13(a)) In this sequence,
only one pulse per 75 ps was injected before the last bit. The measured intensity of
the last output pulse is denominated by A. Second, we assumed that the weakest
output pulse appears in the bit sequence of ‘11111111’. (Fig. 4.13(b)) The measured
intensity of the weakest output pulse is denominated by B. Finally, we obtained
the PIF from A/B. For comparison, ‘0-dominant’ part and ‘1-dominant’ part of thestandard PRBS signal with 27 − 1-bit length are shown in ﬁgs. 4.13(c) and 4.13(d).
We consider that our data-pattern should show more realistic pattern-dependence.
Figure 4.14(a) shows crude estimation results of the output PIF dependence on thedata pattern length for PRBS inputs. We calculated these using the same methodin §3.4.2 for several cw-light intensities (and eﬀective recovery rates). The SOA
injection current, operation frequency, pulse width, and the MZI delay were chosen
to be 400 mA, 200 GHz, 1 ps and 1 ps, respectively. Pattern length of 27 − 1 seems
enough when τeﬀ is shorter than 10 ps. When τeﬀ is around 20 ps, much longer data
pattern with nine successive ‘0’s will be preferable. Since the shortest recovery timeof the SOA B#4 using a holding-beam injection at Iop = 300 mA was about 15 ps(Fig. 4.14(b)), we consider that the data pattern we are using is more suitable thanthe 27 − 1 PRBS data.
The signal extinction ratio was similarly obtained from B/C, where C represents
the signal background level. The background C in our measurements was fairly high.
It was because we could not cut large ASE noise from the EDFA just before thecross-correlator input owing to a short of BPFs. We expect that the extinction ratio
of of the DISC output before ampliﬁcation was much higher.
In the following subsections, we will show systematically-measured results of the
PIF and other properties.

4.3.1

Impact of the holding-beam eﬀect

We ﬁrst studied to how extent the intensity of the cw light aﬀects PIF through theholding-beam eﬀect. Figures 4.15(a) to 4.15(f) show measured output waveforms

83CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(b)

A

0.05

C
0.00
-100

0

’10111010’

0.10’1’ x8
0.05

B
0.00
-100

+50

Time (ps)

(c)
0.15

’0’ x6 ’0’ x5
0.10

0.05

-50

0

+50

Time (ps)(d)
Intensity (arb. unit.)

Intensity (arb. unit.)

-50

0.15

}

’0’ x4 ’0’ x10

}

0.10

SFG Intensity (arb. unit)

0.15

}

SFG Intensity (arb. unit)

(a)

0.15

’00101010’’1’ x7

0.10

0.05

0.00

0.00
-100

-50

0

+50

-100

-50

0

+50

Time (ps)Time (ps)Figure 4.13: Our deﬁnition of the pattern-induced intensity ﬂuctuation (PIF) inthis section. (a) ‘0-dominant’ part of the DISC output waveform. (b): ‘1-dominant’part of the DISC output waveform. PIF is obtained by A/B. (c) and (d): Standard
PRBS signal waveform with 27 − 1-bit length.

84

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(a)
1000

30 ps
20 ps

PIF

100

10 ps
10

τeff = 5 ps
f=200Gb/s

1

3

5

7

9

11

15

2 -1 2 -1 2 -1 2 -1 2 -1

2 -1

(b)

ln (XGM output) (arb. unit)

PRBS pattern length+10

B#4 chip-1

IOP 300 mA

-2
-3
-20

cw input power +11 dBm
pulse 340 fJ
0

+20

+40

+60

Time (ps)Figure 4.14: Crude estimation of the pattern-length eﬀect on the PIF. (a) calculated
PIF dependence on the length of the input PRBS data pattern. (b): measured
example of the gain-recovery proﬁle of the SOA B#4 (τeﬀ ∼ 15 ps).

85

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

and spectra at three cw powers of −10.1 dBm, +3.8 dBm, and +11.2 dBm. The
injection current Iop was 300 mA, and the control pulse energy into the chip was 7.7
fJ. The amount of the nonlinear phase shift was estimated to be 0.2π (for +11.2dBm cw input). The inset of each ﬁgure shows the eye-diagram-like expression,
each containing 32 bits. We could suppress the PIF down to 2.7, using as high cw
power as +11.2 dBm (ﬁg. 4.15(e)). The increase of sub-pulse intensity, which we
predicted in Chapter 3, seemed to cause broadening of the output pulses to ∼ 3.4
ps. Honestly speaking, the impact of the sub-pulses generation seemed somewhat
diﬀerent from what we had expected through Chapter 3. This probably resulted
from large contributions of the carrier-heating related phenomena, which were intentionally suppressed in the previous measurement. Since we could still distinguisheach bit from the output waveform, the result of this eﬀect was not serious in thiscase. Figures 4.16(a) and 4.16(b) show the dependences of the PIF and the average
output power from the DISC on the cw input power, at Iop = 300 mA, 200 mA and
150 mA. We see that the impact of the cw power is extremely large. This explain
why as strong cw power as +16 dBm (into module) was required in the past 168Gb/s wavelength conversion. When the injection current is 200 mA, output power
was 4-dB lower than that for Iop = 300 mA, and when the injection current is 150
mA, output power was 10-dB lower than that for Iop = 300 mA.

4.3.2

Impact of the ﬁlter detuning

Unless we also use other eﬀects to suppress PIF than the holding-beam eﬀect, wehave to consume considerable electric power for the holding-beam ampliﬁcation in
the ultrafast optical gating. Therefore we investigated such eﬀects.
Recently, some researchers have come to utilize detuning of the center wavelengthof the bandpass ﬁlter in ultrafast gating. This ﬁlter-detuning technique was theoretically proposed by Nielsen et. al.[40] Our collaboration team studied the systematicchange of the waveform on the detuning in 2005, and explained the measured patterning to a certain degree[42, 43]. The operating condition of this experiment was,however, fairly limited in some senses.
1. The frequency and the pulse width of the optical signal were only 43 Gb/s
and 2 ps in ref. [42], respectively. Those in ref. [43] were 105 Gb/s and 1.8 ps,
respectively. More contribution of ultrafast phenomena in the SOA should beobserved at higher frequency.

86CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(b)

0.2

0.0
-60

-20

-40

0

Time (ps)

(c)
0.15

PIF=6.2

0.05

-40

-20

0

PIF=2.7

0.06
0.04
0.02
-40

-20

-30
-40
-50
1535

1540

1545

1550

Wavelength (nm)

0
-10
-20
-30
-40
-50
1535

1540

1545

1550

Wavelength (nm)

(f)

0.10

0.00
-60

-20

-60
1530

+20

Time (ps)

0.08

-10

(d)

0.10

0.00
-60

0

-60
1530

+20

Intensity (dBm/nm)

0.1

PIF=36.9(e)
SFG Intensity (arb. unit)

Intensity (dBm/nm)

0.3

Intensity (dBm/nm)

SFG Intensity (arb. unit)

SFG Intensity (arb. unit)

(a)

0

+20

0
-10
-20
-30
-40
-50
-60
1530

1535

1540

1545

1550

Wavelength (nm)Time (ps)Figure 4.15: DISC output signals on several holding-beam intensity. Solid anddashed curves show the ‘1’-dominant and ‘0’-dominant part of the waveform. (a):−10.1 dBm, (b): +3.8 dBm, (c): +11.2 dBm. The insets show eye-diagram-like
expressions. (d) to (f): corresponding optical spectra.

87

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(a)
100

PIF

30

Injection current Iop
300 mA
200 mA
150 mA

10
3

Average output power (dBm)

control pulse energy
7.7 fJ (chip)1-15 -10 -5
0 +5 +10 +15
cw-light power into chip (dBm)

(b)
-10
-15

control pulse energy
7.7 fJ (chip)

-20
-25
-30

Injection current Iop
300 mA
200 mA
150 mA

-35-15 -10 -5 0 +5 +10 +15
cw-light power into chip (dBm)

Figure 4.16: Systematic results on the impact of the holding-beam injection. (a):Dependence of the PIF on the holding-beam intensity, at Iop = 300 mA (solid)
and 200 mA (dashed). (b): Dependence of the average output power on the theholding-beam intensity.

88CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

2. The discussion in ref. [43] is focused on ‘SOA+detuned ﬁlter’ gate rather
than the standard DISC. (SOA+MZI+detuned ﬁlter). Their characteristics
are similar to each other, but not identical. The latter one has been used more
widely.
3. The adopted bit-patterns were ‘0011’ in ref. [42] and ‘0000001111’ in ref. [43],only.
4. Strong holding-beam was not used, and hence two ‘1111’ might not be separated in ref. [43].
Similar and advanced approach has been also investigated [44], but the operation
frequency has been still limited to 40 Gb/s.
Apart from our fundamental study, Liu et. al. demonstrated error-free 160-Gb/s
and 320-Gb/s wavelength conversion using this technique[25, 26]. They succeededin the wavelength conversion of 27 − 1-bit PRBS signal, with much lower cw power
than that used in ref. [23]. Systematic, detailed waveform information including
PIF, sub-pulse generation and so on, however, has not been reported.
Thus we studied the impact of the ﬁlter detuning on the output waveform, inpractical operation conditions. In view of the result of §4.3.1, we operated thegate on two conditions: with strong (+11.2 dBm ) cw power and moderate (+3.8
dBm) cw power. We detuned the 9.2-nm bandpass ﬁlters, one just after the MZI
and another after the EDFA, from the wavelength of the input cw-light (1540 nm).
Figures 4.17(a) and (b) show the output waveforms and spectra on several detuning
conditions, with +11.2-dBm cw-light power. When the ﬁlters were detuned to theblue side by 4.8 nm, the PIF was suppressed from 2.7 to 1.6. In this case, thecw component was just around the edge of the passband, and diminished by 3.5
dB on each ﬁlter. The decrease of the average output power by this detuning was
only 1.8 dB. Farther detuning lead to sub-pulse generation and output degradation,and hence not advantageous. While detuning to the blue side did not distort the
waveform much, detuning to the red side caused complicated waveform distortion.
It was surprising that only 0.7-nm detuning caused large sub-pulses as shown in theﬁgure. When the detuning to the red side reached 4.2 nm, the waveform distortion
disappeared and slight improvement of the PIF was observed. This eﬀect will not
be as useful as that for the blue shift, since it caused larger decrease of the output
power (4.2 dB) and smaller signal-wavelength conversion. Figures 4.18(a) and (b)show the output waveforms and spectra on several detuning conditions, with +3.8dBm cw-light power. The eﬀect of the ﬁlter detuning on the output waveform was
89

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

quite similar to that in the previous case. Detuned to the blue side by 4.8 nm
caused the suppression of the PIF from 6.6 to 2.2, with 1.6-dB output power loss.
This result indicates that the ﬁlter detuning technique largely contributes to the
reduction of the cw-power requirement, and the total energy consumption. Figures4.19(a), 4.19(b), and 4.19(c) show the dependence of PIF, average output power,
and extinction ratio of the output signal on the ﬁlter detuning. Then the impact of
the ﬁlter detuning on the PIF was demonstrated in more practical, high-frequency
condition than in ref. [42]. The optimum ﬁlter detuning in terms of PIF suppression
was 4.8 nm. In such case, the wavelength of the cw light lies just on the edge of
the BPF transmission window. The optimum detuning in terms of the extinction
ratio was slightly shifted to the red side, and the optimum detuning to minimize bit
errors will be somewhere between them. Figures 4.20(a) to (c) show the dependenceof the PIF on the HB intensity when the BPF is blue-shifted by 4.8 nm. These
results show that required HB power was reduced by ∼ 3 dB compared to the casewithout BPF detuning. In each case the reduction of the output power associatedwith BPF detuning was small (Fig. 4.20(d)).

4.3.3

Impact of the nonlinear polarization rotation

In this section, we show that there is another eﬀective method to suppress PIF
of the DISC. This method is based on the nonlinear polarization rotation (NPR)of the cw light in the SOA[63–67]. This eﬀect has been used for some all-opticalsignal processing at relatively low frequency[68–77], but nobody has suggested that
the NPR can be utilized for much higher optical gating when combined with thedelayed-interference scheme.
In Figs. 4.21(a) to 4.21(b), the original scheme of the DISC operation and thatusing the NPR are shown in contrast. When the polarization of the cw light isaligned to TE (or TM) axis of the SOA at the input facet, each control pulse does
not induce any NPR to the cw light (Fig. 4.21(a)). The polarizer in front of the MZI
acts only for reduction of the ASE from the SOA. The output waveform of the DISC√
is approximately proportional to G(t)G(t − ∆t) cos2 (δ∆Φ(t)/2) (see eq. (3.9) in√§3.2), and PIF is induced mostly by the gain modulation, G(t − ∆t). Actually,
the polarization of the cw light is not aligned to TE in most cases, and it rotates on
every pulse injection. If the polarizer in front of the MZI is properly adjusted, thisNPR should cause instantaneous increase in the transmittance of the polarizer andsuppress PIF (Fig. 4.21(b)).

90

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(b)

(a)
0

0.08

PIF=2.5+4.2 nm

-10
-20

0.06

-30

0.04

-40

0.02

-50

0.00

0

0.08

PIF=2.7+2.6 nm

-10
-20

0.06

-30

0.04
-40

0.02

-50

0.00

PIF=2.7-10

Intensity (dBm/nm)

SFG Intensity (arb. unit)

0.06
0.04
0.02
0.00
0.08

PIF=2.6

0.06
0.04
0.02
0.00
0.03

+0.7 nm

-20
-30
-40
-50
0

0

-0.2 nm

-10

-10

-20

-20

-30

-30

-40

-40

-50

-50
-60

0

PIF=1.6

Transmittance (dB)

0.08

0

-4.8 nm

-10
-20
-30

0.02

-40

0.01

-50

0.00

0

PIF=1.8-6.6 nm

-10
-20

0.02

-30
-40

0.01

-50

0.00
-20

0

+20

+40

Time (ps)

+60

-60
1530

1535

1540

1545

1550

Wavelength (nm)Figure 4.17: DISC output signals on several BPF detunings, with strong (+11.2
dBm) holding-beam injection. (a): Monitored waveforms. Solid and dashed curves
show the ‘1’-dominant and ‘0’-dominant part of the waveform. (b): Optical spectra.
The dashed line shows the transmittance spectrum of the 9.2-nm BPF.
91

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING(a)0

PIF=4.6(b)+4.2 nm

-10
-20
-30

0.10

-40
-50
0

0.00

+2.6 nm-10

PIF=10.6

-20
-30

0.10

-40
-50
0

0.00

PIF=7.6

-20

0.05

0.00

PIF=6.6
0.10

0.05

0.00

PIF=4.1

-30
-40
-50
0

0

-0.2 nm

-10

-20

-30

-30

-40

-40

-50

-50

0

-60

-2.9 nm

-10

0.10

-10

-20

Transmittance (dB)Intensity (dBm/nm)

0.10

SFG Intensity (arb. unit)+0.7 nm

-10

-20
-30

0.05

-40
-50

0.00
0.04

0

PIF=2.2-4.8 nm

-10
-20

0.03

-30

0.02

-40

0.01
0.00
-20

-50
0

+20

+40

Time (ps)

+60

-60
1530

1535

1540

1545

1550

Wavelength (nm)Figure 4.18: DISC output signals on several BPF detunings, with moderate (+3.8
dBm) holding-beam injection. (a): Monitored waveforms. Solid and dashed curves
show the ‘1’-dominant and ‘0’-dominant part of the waveform. (b): Optical spectra.
The dashed line shows the transmittance spectrum of the 9.2-nm BPF.
92

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING(a)30

cw-input power into chip
+3.8 dBm+11.2 dBm

Injection current Iop = 300 mA
control pulse energy
7.7 fJ (chip)

10

PIF

Passband of the filters
9.2 nmwavelength separation
between cw and pulse= +15 nm

3

1
-10

-5

0

+5

+10

Average output power (dBm)

Detuning of the BPFs from cw wavelength (nm)

-10

(b)
-15
-20
-25
-30

-10

-5

0

+5

+10

Detuning of the BPFs from cw wavelength (nm)Extinction ratio

10

(c)
3
1
0.3
0.1
-10

-5

0

+5

+10

Detuning of the BPFs from cw wavelength (nm)Figure 4.19: Systematic results on the impact of the BPF detuning. (a): Dependenceof the PIF on the BPF detuning, with strong (+11.2 dBm) and moderate (+3.8
dBm) holding-beam injection. (b): Average output power. (c): Extinction ratio.
93

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(a)

(b)

100

30

PIF

PIF

30

100
Iop 300mA
no BPF detuning
BPF detuning -4.8 nm

103control pulse energy
7.7 fJ (chip)1-15 -10 -5
0 +5 +10 +15
cw-light power into chip (dBm)

Average output power (dBm)

1-15 -10 -5
0 +5 +10 +15
cw-light power into chip (dBm)

(c)
100

PIF

10
3

control pulse energy
7.7 fJ (chip)30

Iop 200mA
no BPF detuning
BPF detuning -4.8 nm

Iop 150mA
no BPF detuning
BPF detuning -4.8 nm

103control pulse energy
7.7 fJ (chip)1-15 -10 -5
0 +5 +10 +15
cw-light power into chip (dBm)

(d)
-10

control pulse energy
7.7 fJ (chip)Iop
300 mA-15
200 mA

-20
-25
-30

150 mAno detuning
BPF -4.8 nm-35-15 -10 -5 0 +5 +10 +15
cw-light power into chip (dBm)

Figure 4.20: Reduction of HB power for PIF suppression by the BPF detuning. (a) to(c): PIF dependence on the HB intensity with and without BPF, at Iop = 150 ∼ 300
mA. (d): Dependence of the average output power on the HB intensity, with andwithout BPF detunings.

94

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING(a)gain G(t)phase Φ(t)MZI

cw light

G(t)G(t-∆t) cos2(δ∆Φ(t))

Polarization(TE polarization)
Pol ( )
Polarizer
Transmittance

SOA∆T

control pulses(b)gain G(t)MZI
cw light

Polarizationcompensate(off TE)

Pol (
SOA

control pulses()

rotatednonlinear polarization
rotation)∆T

Polarizer
Transmittance

Figure 4.21: Schematic images of the suppression mechanism of PIF by NPR in the
DISC. (a): case with no NPR. (b): case when NPR suppresses PIF.

95

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

We demonstrated that such suppression of PIF is actually possible. Figures4.22(a) to (d) show the waveforms of the DISC output, with diﬀerent polarization
settings. We aligned the polarization of the cw light (and that of the control pulses)
at the input facet of the SOA to +30o from its TE axis, using the polarizationmaintaining lensed ﬁber. We also aligned the polarization at P1 to its axis, using
the wave-plates Q1 and H1 . The cw-input power, control pulse energy and injection
current to the SOA were same as those used for the measurement in Figs. 4.18, andno detuning was given to the BPF. The PIF in this case was 5.1. After that, wethe H1 . As the rotation angle ∆θH1 approached to +42o , we observed that the PIF
was drastically reduced down to 1.5, as shown in Fig. 4.22(b) to 4.22(c). The DISC
output signal was almost extinguished around ∆θH1 = 45o , and the PIF increased
again on farther wave-plate rotation (Fig. 4.22(d)). We systematically measured
the dependence of the PIF on ∆θH1 , at several input polarization to the SOA.
The results are shown in Fig. 4.23(a). ∆θH1 was deﬁned separately for each input
polarization series, so that the cw-light without optical modulation went through P1
most at ∆θH1 = 0o . When the input polarization was TE or TM, we could not see
signiﬁcant change of the PIF. When the input polarization was +30o oﬀ from theTE, in contrast, the PIF was reduced down to 1.5 ∼ 3 around +32 ∼ 42o . Similar
reduction of the PIF was obtained when the input polarization was +45o oﬀ fromthe TE, though within narrower ∆θH1 range. We regard these results as a proof of
PIF reduction caused by NPR.
Thus we demonstrated a new method to suppress PIF, in addition to the useof the holding beam and ﬁlter detuning. This method will be as important for thepractical application of the DISC as the ﬁlter detuning technique, since it requires
no additional device from the original conﬁguration. Unfortunately, this method
alone will not provide a perfect solution for the wavelength conversion due to somedrawbacks. One of such drawbacks is the large reduction of the output power, as
shown in Fig. 4.23(b), compared to that for the BPF detuning method. When
we obtained the waveform in Fig. 4.22(c), the average output power decreased
from −15.3 dBm to −26.4 dBm. This decrease is also apparent in the optical
spectra, shown in Figs. 4.22(e) to 4.22(h). Nonetheless, the output waveform in
4.22(c) indicates that its signal-to-noise ratio was still good. When we optimized
the PIF, the extinction ratio was not seriously degraded (4.23(c)). Therefore thismethod should be used together with the other methods, in order to optimize theperformance of the DISC. For that, farther systematic measurements and theoreticalunderstanding of this eﬀect will be quite valuable.
96

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

0.20(a) PIF=5.1 ∆θH1 0o

-10

(e)

-20
-30

0.1

-40
-50
-60
0

0
0.2

(b) PIF=2.9∆θH1 +32o

-10
-20
-30

0.1

-40

Intensity (dBm/nm)

SFG Intensity (arb. unit)

(f)

0
0.08

(c) PIF=1.5∆θH1 +42o

0.06
0.04
0.02

-60
0
-10

(g)

-20
-30
-40
-50
-60
0

0
0.08

-50

(d) PIF=4.2∆θH1 +52o

-10

(h)

-20

0.06

-30
0.04

-40

0.02
0

-50

-50

0

+50

-60
1530

1535

1540

1545

Wavelength (nm)Time (ps)Figure 4.22: Examples of the waveforms and spectra of the DISC output with
suppression of the PIF by NPR. Input polarizations for the SOA are aligned to
TE+30o . (a) to (d): Waveforms with diﬀerent half-waveplate angles. Solid anddashed curves show the ‘1’-dominant and ‘0’-dominant part of the waveform. (e) to(h): corresponding optical spectra.
97

1550

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

(a)

PIF

10

4
SOA input polarization
TE
TE + 30oTE + 45oTE + 90o

2
1

0

+15

+30 +45

+60

+75

+90

+75

+90

Average output power (dBm)

Waveplate rotation angle, ∆θH1 (deg)
-10

(b)

-15
-20
-25
-30
-35

0

+15

+30 +45+60

Waveplate rotation angle, ∆θH1 (deg)10Extinction ratio

SOA input polarization
TE
TE + 30oTE + 45oTE + 90o(c)

3
1
SOA input polarization
TE
TE + 30oTE + 45oTE + 90o

0.3
0.1
0

+15

+30 +45

+60

+75

+90

Waveplate rotation angle, ∆θH1 (deg)Figure 4.23: Dependence of the DISC output properties on the half-waveplate angle
setting. (a): PIF. (b): Average output power. (c): Extinction ratio.

98

CHAPTER 4. 200-GB/S WAVELENGTH CONVERSION USING DISC WITH
LONG-PATTERN WAVEFORM MONITORING

4.4

Conclusion

We performed 200-Gb/s wavelength conversion of long-bit-pattern optical signal
using a DISC, and searched for the requirement on its operating condition to reduce
the output waveform degradation. Signal-waveform monitoring was realized using
the cross-correlation method and the specially-developed ultrafast (∼ 600 fs), lowrepetition rate (40 MHz), low-timing jitter, synchronized sampling pulse generator.
The time resolution of our monitoring system was about 1.5 ps. We measured
the PIF of the DISC output waveform for a practical data sequence, and studied
the impact of the holding-beam injection and BPF detuning on its suppression.
To the author’s knowledge, this is the ﬁrst systematic report on the impact of BPF
detuning in the frequency range over 160 Gb/s. We conﬁrmed that the BPF detuning
suppressed the PIF of the DISC at 200 Gb/s, in a similar way to that in the lower
frequency case (43 Gb/s)[42].
As long as we use only holding-beam eﬀect, approximately 300 mA (580 mW) dc
power consumption and fairly strong holding-beam power (∼ +11.2 dBm into the
SOA chip) were required to reduce the PIF down to 2.7. This power-consumption
value is reasonably close to that calculated for 160-Gb/s operation of the SOA gatesin Chapter 2. When we used detuned BPFs after the SOA, we could suppress PIF
with ∼ 3 dB less HB power. Therefore BPF-detuning technique is promising to
reduce the total power consumption of the SOA-based optical gate.
In addition to these two methods, we also discovered that another method using the nonlinear polarization rotation in the SOA is available to reduce PIF. We
demonstrated that this technique is available when the input polarization to the
SOA is 30 ∼ 45o oﬀ from TE axis, and suppression of the PIF down to 1.5 (from 5)
could be achieved with no additional power consumption through precise polarization control. The required accuracy of the polarization control will be of the order of∼ 5o . We believe that this new method should also contribute to reduce the power
consumption of SOA-based future all-optical gates. Thus farther study on this eﬀect
will be quite worthful.

99CHAPTER 5. CONCLUSIONS AND OUTLOOK

Chapter 5Conclusions and outlook

This chapter gives a summary of the main conclusions of this thesis and takes a lookat the prospective contributions to the future ultrafast all-optical gating research.

5.1

Conclusions of this thesis

The ultrafast all-optical signal processors based on SOAs have been expected as
key elements in future digital networks for a long time, but the lack of their fundamental understanding in many aspects have often caused diﬃculty in their practicaloperations and system designs. Especially, minimum requirements on the operating
conditions such as the electric power consumption have not been clariﬁed so far. In
this thesis, therefore, we aimed to characterize and model the dynamics of SOAs
and SOA-based optical gates to facilitate their applications.
We discussed a new model in Chapter 2, which determines the required amount ofelectrical power consumption of SOAs in XPM-based all-optical gates as a functionof the gate operation frequency B (10 ∼ 160 GHz). In this model, three diﬀerent
carrier-conversion eﬃciencies in the SOA were reasonably introduced to account forthe diﬀerent gain-saturation characteristics against low-frequency optical signals andultrafast optical signals. We predicted that the required amount of the power consumption increases as B 2 (or even faster, depending on the decrease in the conversion
eﬃciencies and diﬀerential gain) and decreases with the total conversion eﬃciency
ηT as ηT−2 in the high-frequency limit. From the systematically-measured results
for the actual SOA samples with diﬀerent structures, we revealed that long SOAs
with an MQW structure had high conversion eﬃciency at high injection current
and required low power consumption. The estimated power consumption value forour best SOA (B#1) was ∼ 750 mW at 160 GHz. We performed the experimental
100

CHAPTER 5. CONCLUSIONS AND OUTLOOK

veriﬁcation of the power consumption calculation and obtained some agreement in
the frequency range up to 40 GHz, though farther need of improvement for higher
frequency region. Thus we made a ﬁrst step to predict, design, and improve theelectric power consumption in the SOA gates. We believe that optimizations of
SOAs for arbitrary operation frequencies in various optical gates become possible infuture with more comprehensive understanding of the conversion eﬃciencies.
In Chapter 3, we discussed the waveform degradation issue regarding DISCtype wavelength converters due to sub-pulse generation originated from exponential
recovery proﬁles of the SOA carriers. This issue might have appeared in the pastgate experiments and have made the gate operation diﬃcult. We precisely observed
and theoretically characterized such generation for the ﬁrst time, in the frequency
range of 10 ∼ 25 GHz using the cross-correlation method. Measured sub-pulse
properties on the relative intensity, waveform dependence on the MZI phase bias,
and intensity increase with the SOA carrier recovery rate were explained using our
DISC-gate model quite accurately. Thus we concluded our present gate model to bea powerful tool to cope with this issue, at least in the frequency range where carrierheating related phenomena do not appear much. Based on this model, we predicted
the trade-oﬀ between the sub-pulse and the patter-induced intensity ﬂuctuation.
Though this issue has not been solved at this moment, we believe that farther
advance in the gate structure supported by our gate-model simulation will provide
a fundamental solution.
In Chapter 4, we studied the waveform degradation, mostly the pattern-induced
intensity ﬂuctuation (PIF), of the output signal from the DISC operated with 200Gb/s, long-bit-pattern (4992 bits) input signal and several methods to suppress
it. In this study, we used our original waveform-monitoring system for the longpatterned signal. When we did not detune the BPF after the SOA (and did not use
the nonlinear polarization rotation), fairly strong holding-beam injection and the
dc current injection of ∼ 300 mA (580 mW) or more was inevitable as estimatedin Chapter 2. Then we systematically measured the impact of BPF detuning, andfundamentally conﬁrmed that it eﬀectively suppressed the PIF of the DISC even at
200 Gb/s and for longer bit pattern (than 27 − 1 , used in ref. [23], [25] and [26]), ina similar way to those in the lower frequency (43 Gb/s), very short-bit-pattern case[42]. This measurement enabled us to evaluate the capacity of the BPF detuning to
mitigate the required holding-beam power (∼ 3 dB). Farther study will contributeto the designing of DISC operation condition with low power consumption. We also
discovered that another method using the nonlinear polarization rotation (NPR) in
101

CHAPTER 5. CONCLUSIONS AND OUTLOOK

the SOA is available to suppress PIF. Suppression of the PIF from 5 down to 1.5
could be achieved with no additional power consumption. We clariﬁed the way tosuppress PIF using this method. This method will be possible with polarizationcontrol accuracy of ∼ 5o . We believe that this new method should also contributeto reduce the power consumption of SOA-based future all-optical gates.
Through these activities, we developed the understanding of the SOA gate
(DISC) operated in the frequency range of 10 ∼ 200 GHz, and considerably facilitated its low-power, small signal-degradation operation and the system design.

5.2

Future outlook

Several kinds of all-optical signal processing have been investigated widely to realize low power, ultrafast signal processing. It has been widely approved that those
based on SOAs are in the nearest position to the practical use. Much potentiallyfaster nonlinear devices as quantum-dot SOAs[82–85], photonic crystal/quantumdot waveguides[86], coupled quantum-well waveguides utilizing intersubband transition (ISBT)[87–90] and so on have been also expected and intensively studied,however, for later use. The future of the SOA gates and other gates is, therefore,
an interesting topic for us.
In Chapter 2, we calculated the dc power consumption in SOA gates. We roughly
deduced that power consumption increases with B 2 . This means that the energy
consumption per one data-bit increases with B, as shown in Fig. 5.1(a). Meanwhile, those of the other optical gates obey diﬀerent relations. For optical gatesusing highly-nonlinear ﬁbers, the only but large source of the power consumption is
the generation of high-intensity optical signals. Since the nonlinear eﬀect in ﬁbers
depends on the peak intensity of the optical signal, the average power of the signal
required for the gating depends rather on the duty ratio than B. This means thatthe energy consumption of the ﬁber gates per one data-bit is inversely proportional
to B (as long as we do not shorten the ﬁber length), in contrast to that for the SOA
gates. We show the estimated power consumption of ﬁber gates also in Fig. 5.1(a).
Here the required signal (and probe) power is assumed to be +29 dBm (and +18
dBm) at 80 ∼ 160 GHz (ref. [16, 17], HNLF length = 1 m, nonlinearity γ = 1100
W−1 km−1 ). The dc electric power consumption value to generate this optical signal
is conservatively estimated to be 8.6 W, in view of the power conversion eﬃciency(∼ 10%) of a typical EDFA (Fig. 5.1(b)). Then the estimated power consumption curves for both kinds of gates cross around 600 GHz. SOA-gates seem to keep102

CHAPTER 5. CONCLUSIONS AND OUTLOOK

(b)

10n
1n

HNLF
ISBT

100p
10p
1p

SOA(B#1)
1G 10G 100G 1T

10T

Bandwidth, B (Hz)Output optical power (mW)

Energy consumption
per bit, (J/bit)

(a)

300

200

100

0

0

1000

2000

3000

DC power consumption in EDFA (mW)

Figure 5.1: Comparison of the power consumption in all-optical gates, and an EDFA
property. (a): Energy consumption per one data-bit against operation frequency.
(b): Typical dependence of EDFA output power on the dc power consumption.
advantages for a while, but the power consumption in SOAs is already substantial
and should be seriously considered. Our result will motivate the researches on otherpassive ultrafast nonlinear devices, which require constant energy per one bit in thegating. For comparison, the estimated energy consumption of the wavelength converter based on the ISBT in the InGaAs/AlAs/AlAsSb coupled quantum wells[87]
was also plotted in the ﬁgure. So far the required power consumption is fairly highcompared to SOA gates or ﬁber gates. If reduction of the power consumption by
factor 10 can be achieved, this gate will be competitive enough.
For the activities to realize much faster SOAs or active nonlinear devices, ourresult suggests certain review. So far most studies on the optimization of the SOA
structure have aimed for the increase of the recovery rate[39, 48, 91, 92]. Our conclusion based on eq. (2.15) is, however, that the increase in the recovery rate itself does
not contribute to reduce the power consumption∗) . We expect that more eﬀorts willbe spent on the improvement of the SOA power consumption eﬃciency after our
research.
Except for Chapter 2, we focused only on the DISC-type wavelength converters.
Nonetheless we consider that our success in the accurate modeling of the gate output
waveform is quite valuable for researchers of other SOA gates. It is true that there
have been plenty of numerical simulation researches for many kinds of SOA gates.
Some researches employed much complexed models in the calculation. The present∗)

Roughly speaking, an SOA with a fast recovery time and also a high gain is likely to require
low power consumption.

103

CHAPTER 5. CONCLUSIONS AND OUTLOOK

author has an impression, however, that most calculated results have not been accurately veriﬁed. For some cases, the present author expect that much improved
results can be obtained by applying our SOA-gate model with properly-estimatedcarrier density values. The most interesting application will be the analysis of 2R
or 3R signal regeneration[56] over 40 GHz.
The all-optical switching devices have not been widely commercialized. For the
wavelength converter or the signal regenerator (2R), so far only a few companies
have produced integrated SMZ devices with guaranteed operation frequencies of up
to 40 GHz) . It is partially because the industrial needs have not been arisen for theimmaturity of some other gate functions as optical memories, and partially becauseof the diﬃculty in the application of these gates themselves. We are conﬁdent
that our research reduced the diﬃculties in realizing much faster (∼ 160 GHz)or much low-power commercial switching devices. For the ultrafast wavelength
converters, integration of the cw-light sources with proper polarization alignment
will be eﬀective. This will support the future realization of large capacity OTDMWDM mesh networks and associated systems (wireless systems beyond 3G, etc.).

)

Examples are: Alphion. Inc., QLightTM ISM, Center for Integrated Photonics Inc., 40G-2RORP, etc.

104

APPENDIX A. ENTIRE WAVEFORM OF THE 200-GB/S INPUT SIGNAL

Appendix A
Entire waveform of the 200-Gb/s input signal

105

APPENDIX A. ENTIRE WAVEFORM OF THE 200-GB/S INPUT SIGNAL

106

APPENDIX A. ENTIRE WAVEFORM OF THE 200-GB/S INPUT SIGNAL

107

APPENDIX A. ENTIRE WAVEFORM OF THE 200-GB/S INPUT SIGNAL108

APPENDIX A. ENTIRE WAVEFORM OF THE 200-GB/S INPUT SIGNAL

whole period109

Acknowledgments

Acknowledgments
I am deeply indebted to Prof. Yoshiyasu Ueno for his supervision over 4 years.
His instruction has been a great help for both my research and my life plan.
I would like to thank the present and former members of Ueno laboratory. Mr.
Kazuki Yanagisawa largely contributed to construct our hand-made OTDM MUX,
which was indispensable for the generation of 200 Gb/s pseudorandom optical signal.
Mr. Masatoshi Okuhira, Mr. Yuichi Inatomi, and Mr. Takeru Yamaji contributed
to fabricate several kinds of hand-made EDFAs. I owe many to Mr. Takehiro
Nishida for supplemental measurements. Dr. Ferran Salleras Villa, Mr. Rei Suzuki,
Mr. Masashi Toyoda, Mr. Ryoichi Nakamoto, Mr. Takashi Ohira, Mr. Masanori
Homma, Mr. Hiroyuki Takeuchi, and other undergraduate students have encouraged
me for a long time. I would like to thank the collaborator; Dr. Kohsuke Nishimura
and Mr. Tomonori Yazaki in KDDI laboratories, Dr. Mads Lønstrup Nielsen, andDr. Motoharu Matsuura in Kishi Laboratory, UEC.
I am quite grateful to Prof. Takeshi Kamiya, Prof. Kenichi Ueda, Prof. Tsuyoshi
Okuno, Prof. Naoto Kishi, and Prof. Kouichi Yamaguchi, who kindly and patiently
took care of the whole examination of Ph.D. degree. I feel that this examination wasa precious opportunity to extend my scope of view. I would also like to thank theexternal reviewers: Prof. Harm J. S. Dorren in Eindhoven University of Technology,
Prof. Jesper Mørk in Technical University of Denmark, and Prof. Jürg Leuthold in
University of Karlsruhe. The positive review comments from the worldwide leading
researchers encouraged me to a great deal. I hope this work provides some assistancefor their researches.
I am deeply indebted to the program leader Prof. Kohzo Hakuta and other staﬀ
of the 21st-Century University’s Center of Excellence (COE) program, “Innovationin coherent optical science” for their invaluable support during my Ph.D. course.
The present work is supported by research project grant #B15360027, “Optical logic gates in the sub-terahertz frequency region,” of the Ministry of Education(MEXT), the research project grant, “Ultra-low-power and ultrafast optical mem110

Acknowledgments

ory/switching device,” of the New Energy and Industrial Technology Development
Organization (NEDO), and 21st Century COE program.
In the last sentence, I would like to thank my parents, who have patiently supported me for a long time.

111

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122

List of publications

List of publications
 Journal papers related to this thesis

1. J. Sakaguchi, M.L. Nielsen, T. Ohira, R. Suzuki, and Y. Ueno, “Observation of small sub-pulses out of the delayed-interference signal-wavelength
converter,” Jpn. J. Appl. Phys. vol. 44, no. 44, pp. L1358-1360, Oct.
2005.
2. J. Sakaguchi, F. Salleras, K. Nishimura, and Y. Ueno, “Frequency-dependent
electric dc power consumption model including quantum-conversion eﬃciencies in ultrafast all-optical semiconductor gates,” Optics Express vol.
15, no. 22, pp. 14887-14900, Oct. 2007.
3. J. Sakaguchi, M. L. Nielsen, T. Ohira, R. Suzuki, and Y. Ueno, “Analysis of Subpulse Generation from Delayed-Interference Signal-Wavelength
Converter for Wide Carrier Recovery Rate Range,” Jpn. J. Appl. Phys.
vol. 47, no. 9, pp. 7182-7189, Sep. 2008.
4. J. Sakaguchi, T. Nishida, and Y. Ueno, “200-Gb/s wavelength conversion
using a delayed-interference all-optical semiconductor gate assisted by
nonlinear polarization rotation,” submitted to Optics Communications.
 Journal papers (co-author)

1. M.L. Nielsen, J. Mørk, R. Suzuki, J. Sakaguchi, and Y. Ueno, “Theoretical and experimental study of fundamental diﬀerences in the noise
suppression of high-speed SOA-based all-optical switches,” Optics Express vol. 13, no. 13, pp. 5080-5086, June 2005.
2. M.L. Nielsen, J. Mork, R. Suzuki, J. Sakaguchi, and Y. Ueno, “Experimental and theoretical investigation of the impact of ultra-fast carrier
dynamics on high-speed SOA-based all-optical switches,” Optics Express,
vol. 14, no. 1, pp. 331-347, Jan. 2006.

123

List of publications

3. Y. Ueno, R. Nakamoto, J. Sakaguchi, and R. Suzuki, “Optical-spectrumsynthesizer design within an all-optical semiconductor gate to reduce
waveform distortion induced by carrier-cooling relaxation at sub-Teraherz
frequencies,” Optics Express vol. 14, no. 26, pp. 12655-12664, Dec 25,
2006.

 Contributions to international conferences (ﬁrst author)

1. J. Sakaguchi, M.L. Nielsen, T. Ohira, R. Suzuki, and Y. Ueno, “Observation of small sub-pulses generated in the DISC-type wavelength converter,” 10th Optoelectronics and Communications Conference (OECC2005), Seoul, Korea, July 5-8, 2005, paper 8F1-4.
2. J. Sakaguchi, Y. Ueno, K. Nishimura, and T. Yazaki, “New method forcharacterizing the injected-carrier-to-photon conversion eﬃciencies insidethe ultrafast all-optical semiconductor gates,” 9th International Symposium on Contemporary Photonics Technology (CPT 2006), Tokyo, Jan.
11-13, 2006, paper no. P-26, pp. 130-131.
3. J. Sakaguchi, Y. Ueno, K. Nishimura, and T. Yazaki, “New method forcharacterizing the injected-carrier-to-photon conversion eﬃciencies insidethe ultrafast all-optical semiconductor gates,” Optical Ampliﬁers andtheir Applications (OAA 2006), June 25-28, 2006, Whisler, Canada, paper no. OTuC4.
4. J. Sakaguchi, F. Salleras, and Y. Ueno, “Frequency-dependent electricdc power consumption in ultrafast all-optical semiconductor gates,” 12th
Optoelectronics and Communications Conference (OECC 2007), July 913, 2007, Yokohama, Japan, paper no. 10D2-3.

 Contributions to international conferences (co-author)

1. M.L. Nielsen, J. Mork, J. Sakaguchi, R. Suzuki, and Y. Ueno, “Reductionof nonlinear patterning eﬀects in SOA-based all-optical switches using

124

List of publications

optical ﬁltering”, Optical Fiber Communication Conference (OFC 2005),
Los Angeles, March 6-11, 2005, paper OThE7.
2. R. Suzuki, S. Kobayashi, J. Sakaguchi, and Y. Ueno, “Threshold condition for pulse generation from a DISC-loop-type pulse generator,” IQEC/CLEOPR, July 11-15, 2005, Tokyo, paper CMF1-4, pp. 1522-1523.
3. R. Suzuki, T. Ohira, J. Sakaguchi, and Y. Ueno, “40-GHz mode-locked
pulse generation with a new scheme of SOA-based pulse generation,”
CLEO/QELS 2006, May 21-26, 2006, Long beach, USA, paper no. CMG5.
4. F. Salleras, M. Honma, J. Sakaguchi, Y. Ueno, N. Ozaki, Y. Kitagawa,
K. Asakawa, N. Ikeda, Y. Sugimoto, “Criterion for removing a delayed
peak from the ultrafast nonlinear response of photonic crystal/quantum
dot waveguides,” 1st Nonlinear Photonics Topical Meeting (NP 2007),
Optical Society of America, Quebec city, Canada, Sept. 2-6, 2007, paper
no. NTuC6.
5. R. Nakamoto, H. Takeuchi, J. Sakaguchi, and Y. Ueno, “1.55-µm, modelocked, single-longitudinal-mode, 10-GHz, 2-ps, ultra-short optical pulse
train from our original semiconductor-based pulse-source scheme,” abstract submitted to the Topical Conference on Nanophotonics (NANO),
Optical Society of America, May 26-29, 2008, Southeast Univ., Nanjing,
P.R. China. paper no. Nano-08-191.

125