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Paper
Study of Coupling Loop Interference Canceller for SC-FDE Wireless System with High
Order Modulation
Tomoyuki Suzuki †1, Masato Yokoi †2, Hotaka Ogawa †1,
Hiroyuki Hamazumi †3（member） and Tomohiko Kimura †4（member）

Abstract: In wireless transmission for broadcasting services, such as wireless cameras, OFDM, which has excellent multipath resistance, is used.
Orthogonal Frequency Division Multiplexing (Orthogonal Frequency Division Multiplexing) has been widely used, but in recent years, the Single Carrier - Frequency Domain
Equalization (SC-FDE) method has been attracting attention because it combines good power efficiency with multipath resistance through frequency domain equalization. The
booster of the SC-FDE method, which contributes to extending the transmission distance, is a regenerative relay that demodulates and remodulates the received data and retransmits it.
This allows the construction of a Decode and Forward relay (DF) that can achieve relaying without signal degradation. However, in relaying using the same frequency , multipath
and coupling loop interference (CLI) become problems. We have been studying a CLI canceller for the SC-FDE method that can expand the transmission area while efficiently
using the frequency band. However, there was an issue that the steady-state error after the convergence of the cancellation algorithm was large and it could not handle high-ordermodulation. In this paper, we propose a CLI canceller using a Unique Word (UW) that is inserted as a reference signal into the transmission signal of the SC-FDE method, andshow the possibility of constructing a Single Frequency Network (SFN) that supports high-order modulation.

Keywords: SC-FDE, booster, CLI canceller , micrometer waveband, high order modulation

We have investigated a single frequency network (SFN) booster using a coupling loop

1. Introduction
Wireless cameras, which can transmit video and audio wirelessly in the production ofinterference (CLI) canceller and demonstrated the possibility of expanding the coveragearea. In addition, we have demonstrated that a decode and forward relay (DF) can besports broadcasts, music programs, and other such programs, enable highly mobile

configured to demodulate, remodulate, and retransmit the received data, and that relay

program production without the constraints of cables.

transmission without signal degradation is possible for low-order modulations below

In mobile wireless transmission of program material from wireless cameras and other devices, OFDM
(Orthogonal Frequency Division Multiplexing) is used, which has excellent resistance to multipath interference.

QPSK2 ). However, the CLI cancellers proposed so far have a large steady-state error andcannot handle high-order modulations above 16QAM2).

In recent years, however, single carrier-frequency domain equalization (SC-FDE) has
become popular, which combines power efficiency with multipath resistance throughfrequency domain equalization.

Expanding the coverage area using SFN boosters is an effective method not only for

In particular, the SC-FDE method for professional wireless cameras using millimeter wave

millimeter wave bands but also for microwave bands3). In particular, higher-orderbands has been attracting attention.

modulation beyond 64QAM has been standardized using microwave band OFDM4).

It is a valuable piece of filming equipment that can transmit HDTV and UHDTV signals withlow latency.1 ) However, millimeter wave radio waves are subject to high attenuation,
resulting in a narrow coverage area.
The authors have proposed a method for applying SC-FDE to the millimeter wave band.

In this paper, we propose a CLI cancellation algorithm using UW (Unique Word) inserted
in the transmission signal of the SC-FDE system to realize an SFN booster that can handle
high-order modulation beyond 16QAM, and further propose the Double RLS algorithm asa sequential coefficient update algorithm for the CLI canceller.

Received on December 18, 2023, re-accepted on March 12, 2024, and collected on June 6, 2024. †1 Department ofElectrical and Electronic Engineering, Tokyo National College of Technology (1220-2,
Kunugi-cho, Hachioji, 193-0997, Japan, TEL: 042-668-5111) †2 Department ofMathematics, School of Science, Tokyo Institute of Technology

We compared the two algorithms, Recursive Least Square (RLS), and the proposed(2-12-1 Ookayama, Meguro-ku, Tokyo 152-8551, TEL: 03-5734-2205)†3 Department of Electrical and Electronic Engineering, Fukui National College of Technology(Shimojyocho, Sabae City, Fukui Prefecture, 916-8507, Japan, TEL: 0778-62-8256) †4method for SC-FDE wireless systems supporting high-order modulation.
The possibility of constructing a CLI canceller is examined by computer simulation.

Department of Electrical Engineering, Tokyo National College of Technology(1220-2, Kunida-cho, Hachioji-shi, 193-0997, TEL: 042-668-5177)

This will be clarified through the analysis.

(135) 499
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Journal of the Institute of Image Information and Television Engineers, Vol. 78, No. 4, pp. 499-509 (2024)

論 文
高次変調を行う SC-FDE 無線伝送システムのための回り込みキャ
ンセラの検討Study of Coupling Loop Interference Canceller for SC-FDE Wireless System with High Order
Modulation
鈴 木 智 之 †1，

横 井 将 人 †2，
正会員

小 川 穂 高 †1，

濱 住 啓 之 †3，正会員 木 村 知 彦 †4

Tomoyuki Suzuki †1, Masato Yokoi †2, Hotaka Ogawa †1, Hiroyuki Hamazumi †3 and Tomohiko Kimura †4
あらましワイヤレスカメラをはじめとする放送業務用の無線伝送においては，マルチパス耐性に優れる OFDM

（Orthogonal Frequency Division Multiplexing）方式が広く利用されてきたが，近年は電力効率の良さと周波数領域
等化によるマルチパス耐性を併せ持つ SC-FDE（Single Carrier - Frequency Domain Equalization）方式が注目されている．伝送距離の拡大に資する SC-FDE 方式のブースターは，受信データの復調，再変調を行い再送信する再生中継（DF: Decode and Forward relay）を構成でき，信号劣化のない中継が実現できる．しかし，同一周波数によるリ

レー中継では，マルチパスや回り込み波（CLI: Coupling Loop Interference）が問題となる．筆者らは，周波数帯域を有効に利用しつつ伝送エリアを拡大できる SC-FDE 方式に対応する回り込み（CLI）キャンセラを検討してきた．しかしながら，キャンセルアルゴリズムの収束後の定常誤差が大きく，高次の変調に対応できない課題があった．本論文
では，SC-FDE 方式の送信信号に参照信号として挿入する UW（Unique Word）を用いた CLI キャンセラを提案し，高次変調に対応した SFN（Single Frequency Network）構築の可能性を示す．
キーワード： SC-FDE，ブースター，CLI キャンセラ，マイクロ波帯，多値化り込み（CLI: Coupling Loop Interference）キャンセラを用い

１．ま

え

が

き

た SFN（Single Frequency Network）ブースターの検討を行

スポーツ中継や音楽番組などの制作業務において無線でい，カバーエリアを拡大できる可能性があることを示してき

映像や音声を伝送できるワイヤレスカメラは，ケーブルのた．加えて，受信データの復調，再変調を行い再送信する再制約がなく機動性が高い番組制作が実現できる．

生中継（DF: Decode and Forward relay）が構成でき，QPSK

ワイヤレスカメラなどの番組素材の移動無線伝送において以下の低次変調の場合は信号劣化のない中継伝送が実現可能

は，マルチパス耐性に優れるOFDM（Orthogonal Frequency

であることを明らかにしてきた 2）．しかし，これまでに提案

Division Multiplexing）方式が広く利用されてきたが，近年はした CLI キャンセラは定常誤差が大きく，16QAM 以上の高電力効率の良さと周波数領域等化によるマルチパス耐性を併

次の変調には対応できない課題があった 2）．

せ持つ SC-FDE（Single Carrier - Frequency Domain

SFN ブースターによるカバーエリアの拡大は，ミリ波帯

Equalization）方式が注目されている．特に，ミリ波帯を用のみならずマイクロ波帯 3）においても有効な手法である．

いた SC-FDE 方式の業務用ワイヤレスカメラは，高品質な特に，64QAM を超える高次変調は，マイクロ波帯 OFDMHDTV 信号や UHDTV 信号を低遅延で伝送できる貴重な撮による標準化の実績がある 4）．

影機材である 1）．しかし，ミリ波帯の電波は減衰が大きく，カバーエリアが狭くなる課題がある．
筆者らは，ミリ波帯において SC-FDE 方式に適用できる回

本論文では，16QAM を超えるような高次の変調に対応できる SFN ブースターを実現するため，SC-FDE 方式の送信信号に挿入される UW（Unique Word）を用いた CLI キャ
ンセルアルゴリズムを提案し，さらに CLI キャンセラの逐

2023 年 12 月 18 日受付，2024 年 3 月 12 日再受付，2024 年 6 月 6 日採録†1 東京工業高等専門学校 専攻科 電気電子工学専攻（〒 193-0997 八王子市椚田町 1220-2，TEL 042-668-5111）†2 東京工業大学 理学院 数学系（〒 152-8551 目黒区大岡山 2-12-1，TEL 03-5734-2205）†3 福井工業高等専門学校 電気電子工学科（〒 916-8507 福井県鯖江市下司町，TEL 0778-62-8256）†4 東京工業高等専門学校 電気工学科（〒 193-0997 八王子市椚田町 1220-2，TEL 042-668-5177）

映像情報メディア学会誌 Vol. 78, No. 4, pp. 499 〜 509（2024）

次係数更新アルゴリズムとして Double RLS アルゴリズムを提案する．従来の LMS（Least Mean Square），RLS（Recursive Least Square）の両アルゴリズム方式との比較を行い，高次変調に対応した SC-FDE 無線システム用のCLI キャンセラを構築できる可能性を計算機シミュレー
ションにより明らかにする．
（135） 499

Machine Translated by Google
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Because this is a machine translation, it may contain typographical errors, mistranslations, or parts of the paper that have not been reflected in the translation.

In order to suppress noise, interference, and distortion components contained in

2. SC-FDE method

UWr, filter processing (Block filter) is performed for each FFT block after frequency

The SC-FDE system adds a reference signal to a single-carrier transmission
signal and transmits it, and uses the reference signal to perform frequency domain

domain conversion. The filtered UWr (for a block of the size of the first half of UWF
and the second half of UWB combined) isN

equalization at the receiver. The SC-FDE system has a smaller ratio of average

The UW generator (UW gen.) generates

power to peak power than the OFDM system, which allows for a larger amplifier

The transmission channel characteristics are estimated by dividing by UWtrue (CE:output power. Figure 1(a) shows a block diagram of the transmitter section of the

Channel Estimation). The received signal is divided by the estimated transmission

SC-FDE system, and Figure 1(b) shows a block diagram of the receiver section.

channel characteristics (FDE) and filtered with a roll-off filter (NYQ filter) thatIn the transmission section of the SC-FDE system, a UW is inserted as asatisfies the Nyquist criterion. Finally, the signal is converted from the frequency

reference signal into the transmission signal that has been QAM modulated (QAM

domain to the time domain (IFFT) and QAM demodulated (QAM Demo.), making itmod.), and the signal is transmitted through a transmitting filter. The transmission

possible to extract only the original transmitted signal components.

signal structure is as shown in Figure 2. By inserting a UW before and after thesignal block, it is possible to enhance multipath resistance in addition to frequency
domain equalization on the receiving side.
Let N be the block size , M be the data block size , and L be the transmission block

3. Booster
A booster amplifies the received signal and retransmits it at the same frequency.

size. When the data block is frequency-domain transformed (FFT: Fast FourierThis is an easy way to extend the transmission distance, but there are issues withTranslation) on the receiving side,

this method, such as the radio waves emitted from the transmitting antenna being

A block of size M + N is used , combining the latter half of UWF and the first half ofreflected back to the receiving antenna, degrading the transmission characteristics,

UWB . The UW code is the Zadoff-Chu code5) (hereafter referred to as Chu code),and the booster causing oscillation.

which has a constant amplitude and excellent autocorrelation characteristics. The

In addition, there is a multipath in the propagation path from the parent station

same Chu code is used for both UWF and UWB.

to the booster, so it is necessary to achieve both CLI cancellation and multipath

Even if the Chu code is cyclically shifted, the Chu code maintains its continuity andequalization. To solve these problems, a previous study investigated a booster thatits characteristics do not change. Therefore, UWF and UWB are connectedcombines a CLI canceller and time-domain multipath equalization, and demonstrated

continuously.

its effectiveness in QPSK modulation2).

In the receiving section, the received signal is filtered by a receiving filter, andthen the signal is transformed into the frequency domain.
The signal is then subjected to a fast Fourier transform (FFT) and then subjected to frequency domain equalization (FDE) to remove

However, it was not possible to handle high-order modulation.
We investigated a CLI canceller that can handle the modulation.

noise, multipath interference, nonlinear distortion, and other transmission path characteristics that are added during the transmission path.

3.1 Overview of the booster's internal processing

The channel characteristics required for frequency domain equalization are

A schematic diagram of the booster configuration is shown in Figure 3. The booster's

estimated using a known UW.

received signal y(n) is expressed as the sum of the parent station signal s(n), the parent
station signal multipath g(n), the interference signal qH(n)u(n), and noise nAWGN(n) as
shown in equation (1).

(1)
Here, the diffraction wave qH(n)u(n) is the booster output signal u(n) multipliedby the diffraction path characteristic qH(n), and [ ]H is the operator representing theHermite transpose.

Fig. 1 SC-FDE.

Figure 2 Block diagram of SC-FDE.

500 (136)

Fig. 3 Booster with CLI canceller.

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Unauthorized use of original papers or translated papers is prohibited.
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UWr に含まれる雑音・干渉・歪成分を抑圧するため，周波２．SC-FDE 方式

数領域変換後に，FFT ブロックごとにフィルタ処理

SC-FDE 方式は，シングルキャリヤの送信信号に参照信

（Block filter）を行う．フィルタ処理された UWr（UWF の前号を付加して送信し，受信側で参照信号を利用して周波数半と UW B の後半を合わせた N サイズのブロックに対して領域等化を行う方式である．SC-FDE 方式は，OFDM 方式

FFT する）を UW 生成器（UW gen.）により生成されたよりもピーク電力に対する平均電力の比が小さく，このた

UWtrue にて除算することで伝送路特性を推定する（CE:め増幅器の出力電力を大きくできる．図 1（a）に SC-FDE 方

Channel Estimation）．推定した伝送路特性で受信信号を除式の送信部，図 1（b）に受信部のブロック図を示す．

算（FDE）し，ナイキスト基準を満たすロールオフフィル

SC-FDE 方式の送信部において，QAM 変調（QAM mod.）タ（NYQ filter）にてフィルタリングする．最後に，周波数を行った送信信号に参照信号として UW を挿入し（UW

領域から時間領域に変換（IFFT）し，QAM 復調（QAMins.），送信フィルタ（Transmitting filter）を通し送信する．

Demo.）することで，本来の送信信号成分のみを取り出す

送信信号の構造は，図 2 に示すように，信号ブロックの前ことが可能となる．

後に UW を挿入することで，受信側での周波数領域等化に加え，マルチパス耐性を高めることができる．UW のブ

３．ブースター

ロックサイズを N，データブロックサイズを M，送信ブ

ブースターは受信信号を増幅し，同一周波数で再送信を

ロックサイズを L とする．受信側でのデータブロックを周行うものである．伝送距離を簡単に拡大できる手法である

波数領域変換（FFT: Fast Fourier Translation）する際に，が，送信アンテナから放射された電波が受信アンテナに回UW F の後半と UW B の前半を合わせた M + N サイズのブり込み，伝送特性を劣化させたり，ブースターが発振を起ロックを使用する．また UW の符号としては，振幅が一定

こしたりする課題がある．

で自己相関特性に優れる Zadoff-Chu 符号 5）（以後 Chu 符号）

また，親局からブースターまでの伝搬路にはマルチパス

を用い，UW F と UW B ともに同一の Chu 符号を用いる．

が存在し，CLI キャンセルとマルチパス等化を両立させる

Chu 符号は，巡回シフトをしても Chu 符号としての連続性必要がある．これらの課題を解決するために，先行研究でを保ちその特性は変わらない．そのため，UWF と UWB とは，CLI キャンセラと時間領域マルチパス等化を組み合わの間は連続的に接続される．

せたブースターを検討し，QPSK の変調においてその有効受信部では，受信信号を受信フィルタ（Receiving filter）
にてフィルタリングした後，信号を周波数領域に変換（FFT）し，伝送路で付加された雑音やマルチパス干渉，非性を示した 2）．
しかし，高次変調には対応できていなかったため，高次変調に対応する CLI キャンセラの検討を行った．

線形歪みなどの伝送路特性を周波数領域等化（FDE）によ

3.1

り除去する．周波数領域等化に必要な伝送路特性の推定は，

図 3 にブースターの構成の概要図を示す．ブースターの既知の UW を用いる．伝送路を通り受信した受信参照信号

ブースター内部処理の概要

受信信号 y(n)は，親局波 s(n)，親局波マルチパス g(n)，回
り込み波 qH(n)u(n)，雑音 nAWGN(n)の和として（1）式のように表される．
(1)

ここで，回り込み波 q H (n)u(n)はブースター出力信号
u(n)に対して回り込み伝搬路特性 qH(n)を掛けたもの，[ ]H
は Hermite 転置を表す演算子である．

図 1 SC-FDE.

図 2 SC-FDE block diagram.

500 （136）

図 3 Booster with CLI canceller.
映像情報メディア学会誌 Vol. 78, No. 4（2024）

Machine Translated by Google
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The interference qH(n)u(n) and the master station multipath g ( n ) contained inThis will reduce the error in the steady state.

the received signal y(n) are respectively eliminated by the CLI canceller (CLI

3.1.3 Demodulation, decision and remodulation

The interference is removed using a Canceller and a Linear Equalizer, and finally aExtract the data portion from the linearly equalized signal l(n),regenerative repeater mechanism (Re-modulator) is incorporated to demodulate,

The bit is determined by QAM demodulation. The bit-determined signal is then QAM

judge, and re-modulate the signal from the parent station s(n) and retransmit it.

modulated again to restore the parent station signal s(n). Furthermore, the UW true

3.1.1 CLI Canceller The CLI canceller block has anvalue UWtrue(n) regenerated within the booster is added in the same way as the

input u(n), an internal adaptive filter (FIR filter) with a tap length of D and a tap

transmission signal in Figure 2. By performing this demodulation, determination,and re-modulation, it is possible to completely eliminate relay degradation.

coefficient of w(n), which are

3.1.4 Booster Processing Delay The(2)booster output signal u(n) is delayed by the filtering process in the booster. In
the computer simulations in this paper, a filtering delay of 28 samples is assumed(3)
The output signal of the adaptive filter is expressed as wH(n)u(n).

to occur in the Transmitting Filter and the Receiving Filter, respectively, and a totalof 56 samples is set as the Booster delay, which is the delay from reception toequalization processing, CLI cancellation, remodulation, and transmission. The

Moreover, the signal y0(n) after CLI cancellation is

Booster delay value does not affect the linear equalizer in the booster and the loop
(Four)

cancellation performance. However, in the receiver, in areas where both the parent
station wave s(n) and the booster wave u(n) are received, if the delay time differenceThis can be expressed as:

The adaptive algorithm adjusts the tap coefficient w(n) so that the adaptive filter

between the parent station wave and the booster wave exceeds the equalization

output wH(n)u(n) approaches the diffraction grating qH(n)u(n) . If a tap coefficient

range by the UW (corresponding to the length of N in Figure 2) , the frequency

w(n) that satisfies qH(n) = wH(n) can be obtained , the diffraction grating qH(n)u(n)domain equalization performance may deteriorate2).

can be completely cancelled. Note that the diffraction grating propagation pathcharacteristic qH(n) has an infinite degree in real space, but if the main diffraction
grating components can be expressed by the tap coefficient w(n), the diffraction

Other simulation parameters are shown in Tables 1 to 4 of Chapter 4.
The results are

grating can be cancelled sufficiently.
be able to.

shown in Table 3. 3.2 Conventional CLI
Canceller The configuration of the CLI canceller proposed in Reference 2) is

3.1.2 Linear Equalizer Figure 4 shows an overview
of a linear equalizer. For the parent station signal multipath g(n), a linear equalizer
is used to perform equalization using the LMS (Least Mean Square) algorithm. The

shown in Figure 5. To distinguish it from the computer simulation described later,
this method is referred to as the “Data method (Data).”signal yf(n) after the booster receive filter is input, and the algorithm is run to

Tap update calculations are performed for all data blocks of lock length L. Tap

minimize the difference between the demodulated, judged and remodulated signal

weights are updated without distinguishing between UW and Data.

r(n) and the filter output l(n) , and the calculated tap coefficient wl(n) is passed to

In the reference 2), the taps are updated based on the gradient descent method

the FIR filter for filtering . The tap coefficient update formula when the step size is(GD), and the update formula for the tap coefficient w(n) is given by the step size ÿ,µm is expressed as follows:

the signal y0(n) after CLI cancellation, and the complex conjugate u*(n) of u(n) , asfollows:(7)(Five)

This can be expressed as:(6)
The LMS step size µm of the linear equalizer is set small in order to suppress
steady-state errors.

In this paper, we also use a recursive least squares (RLS) algorithm to achieve the conventional method.

RLS (Reduced Least Square) algorithm6) applied to the data
We also prepared the data (Data) and compared it with the proposed method shown in the next section.

Figure 4 Linear Equalizer.

Paper ÿ Study on Coupling Interference Canceller for SC-FDE Wireless Transmission System Using High-Order Modulation

Figure 5 Previously proposed CLI canceller.

(137) 501

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受信信号 y(n)に含まれる回り込み波 qH(n)u(n)，親局波マ
ル チ パ ス g ( n ) は ， そ れ ぞ れ ， C L I キ ャ ン セ ラ（ C L I

落ちるが，定常状態における誤差を減らすことができる．
3.1.3

復調・判定・再変調

Canceller）と線形等化器（Linear Equalizer）により除去し，線形等化後の信号 l(n)から，データ部分を切り出し，

最後に復調・判定・再変調を行う再生中継の機構（Re-QAM 復調することでビット判定する．ビット判定した信

modulator）を組み込み，親局波 s(n)を復元して再送信する．

号を再度 QAM 変調し，親局波 s(n)を復元する．さらに，

3.1.1

CLI キャンセラ（CLI Canceller）

ブースター内で再生した UW 真値の UWtrue(n)を図 2 の送信

CLI キャンセラブロックは，ブロックへの入力を u(n)，内部の適応フィルタ（FIR Filter）のタップ長を D，タップ

信号と同様に付加する．この復調・判定・再変調を行うことで，中継劣化を完全になくすことが可能となる．
3.1.4

係数を w(n)とし，それぞれ，(2)

ブースター処理遅延

ブースター出力信号 u(n)は，ブースター内におけるフィ
ルタリング処理により遅延して出力される．本論文の計算(3)と表され，適応フィルタの出力信号は，wH(n)u(n)となる．
また，CLI キャンセル後の信号 y0(n)は，

機 シ ミ ュ レ ー シ ョ ン で は ， Transmitting Filter とReceiving Filter において，それぞれ，28 sample のフィル
タリング遅延が発生するとし，合計 56 sample を受信から(4)等化処理や CLI キャンセルを行い再変調して送信するまでの遅延 Booster delay と設定した．Booster delay の値は，ブースター内の線形等化器と回り込みキャンセル性能にはと表される．
適応アルゴリズムは，適応フィルタの出力 wH(n)u(n)が，影響は及ぼさない．しかし，受信機においては，親局波

回り込み波 qH(n)u(n)に近づくようタップ係数 w(n)を調整

s(n)とブースター波 u(n)を両方受けるエリアでは，親局波する．qH(n) = wH(n)を満たすタップ係数 w(n)が得られれとブースター波の遅延時間差が UW による等化範囲（図 2，ば，回り込み波 q H(n)u(n)を完全にキャンセルすることがN の長さに相当）を超えると，周波数領域等化性能が低下できる．なお，回り込み伝搬路特性 q H(n)は，実空間上でする場合がある 2）．

無限の次数を持つが，主要な回り込み成分をタップ係数w(n)で表現できれば，充分に回り込み波をキャンセルする

3.2

ことができる．
3.1.2

その他のシミュレーションパラメータは第 4 章の表 1 〜
表 3 に記す．
従来方式 CLI キャンセラ

文献 2）で提案した CLI キャンセラの構成を図 5 に示す．

線形等化器（Linear Equalizer）

図 4 に線形等化器の概要を示す．親局波マルチパス g(n)後述する計算機シミュレーションでの区別のため，こちらに関しては，線形等化器（Linear Equalizer）を用いて LMSの手法を「データ方式（Data）」と表記する．図 2 に示したブ（Least Mean Square）アルゴリズムにより等化を行う．

ロック長 L のデータブロックすべてでタップ更新の計算をブースター受信フィルタ後の信号 yf(n)を入力とし，復調・行う．UW と Data を区別せずタップ重みを更新する．

判定・再変調を行った信号 r(n)とフィルタ出力 l(n)の差を

文献 2）では，最急降下法（GD: Gradient Descent）をベー最小化するようアルゴリズムを動作させ，それにより計算スにしたタップ更新を行っており，タップ係数 w(n)の更新

したタップ係数 wl(n)を FIR Filter に渡しフィルタリングを式は，ステップサイズ µ，CLI キャンセル後信号 y0(n), u(n)行う．ステップサイズを µ m としたときのタップ係数の更の複素共役 u*(n)を用いて，新式は次式のように表される．

(7)
(5)

と表される．
(6)

また，本論文では，従来方式に RLS（Recursive Least

なお，線形等化器のLMS Step size µm については，定常誤Square）アルゴリズム 6）を適用したデータ方式の RLS（RLS

差を抑える観点から小さく設定する．これにより収束速度は（Data））も用意して次節に示す提案手法との比較を行った．

図 4 Linear Equalizer.

図 5 Previously proposed CLI canceller.

論

文 □ 高次変調を行う SC-FDE 無線伝送システムのための回り込みキャンセラの検討（137） 501

Machine Translated by Google
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The tap coefficient update formula by RLS (Data ) is the same as that shown

This is to allow the step size to follow the fluctuations of the CLI, but while thein Section 3.5, where d(n) = y0(n). 3.3 CLI Canceller for High-convergence speed improves, the steady-state error increases.

Order Modulation
The CLI canceller proposed in this paper, which uses only UW for learning,

3.5 RLS Algorithm Evaluationis shown in Figure 6. To distinguish it from the computer simulation described

Function for the RLS Algorithm

J is the forgetting factor ÿ As such,later, we refer to this method as the "UW method."(11)

The CLI cancellation method using UW described in this section is the sameas the channel characteristic estimation method using UW in the frequency
domain equalization (FDE) of the receiver described in Chapter 2.

The RLS algorithm, which incorporates past signals as weights into the

The UWevaluation function, is characterized by its high convergence. The update
formula is as follows:part extracted from the received signal y(n) at the booster is
If UWbr(n) is the UW signal and the true UW value regenerated within the booster is UWtrue(n) ,(12)then the difference between these becomes the interference component d(n).

(8)

(13)

Here, by subtracting UWtrue(n), which corresponds to the parent station wave s(n),(14)

from equation (1), the interference component d(n) becomes a UW consisting only of
the parent station wave multipath g(n), the diffraction wave qH(n)u(n), and the noise(15)

nAWGN(n) component .

The interference component d(n) is input to the adaptive algorithm as a target
value , and the tap coefficient w(n) of the adaptive filter is adjusted so that thewhere k(n) is the gain vector and P(n) is the sum of the correlation matricesof the inputs.

retransmitted signal u(n) approaches d(n) . The adjustment of the adaptive filter(16)

taps is controlled so that it is performed only in the UW portion.

The adaptive algorithm used is the Double RLS algorithm, which uses twostages of the proposed RLS algorithm. The LMS and RLS algorithms6) were

Inverseÿ This is an estimate of –1(n).

used as comparison algorithms. In addition, the GD (Data) method, which

matrix of 3.6 Double RLS algorithm

updates the tap weights without distinguishing between the conventional UW

Figure 7 shows the Double RLS algorithm.

and Data, and the GD (UW) method, which updates the tap coefficients only in

The structure of the (DRLS) algorithm is shown below.

the UW portion, were prepared. The update formula used was the one shownin Section 3.2 for the conventional CLI canceller. 3.4 LMS Algorithm

Two RLS filters are used in parallel, and the output of the first RLS filter, w1
H(n)u(n), uses the interference component d1(n) and the booster output signal
u(n) as input, and is the same as the filter output when a normal RLS algorithm

The update formula for the tap coefficient w(n) using the LMS algorithm is

is used.

described by the following formula using the step size µ, the interference(17)

component d ( n ), the filtered output wH(n)u(n), and the booster output signal
u(n) . Note that the step size and input signal are different from those of the(18)

LMS algorithm used in the linear equalizer.

(19)

(9)
(Ten)

The step size µ of the LMS used in the CLI canceller is a linear equalizer.

Fig. 6 CLI canceller using UW for adaptive algorithm.

502 (138)

Fig. 7 Double RLS algorithm.

Journal of the Institute of Image Information and Television Engineers Vol. 78, No. 4 (2024)【Copyrights to Machine-Translated Content】
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RLS（Data）によるタップ係数更新式は，3.5 節に示す式にに用いる LMS の step size µm よりも大きく設定する．これ

おいて，d(n) = y0(n)とした場合と同じになる．

は，CLI の変動に追従させるためであるが，収束速度は向3.3

高次変調対応 CLI キャンセラ

上する一方，定常誤差が増えることになる．

本論文にて提案する，UW のみを学習に使用する CLI
キャンセラの構成を図 6 に示す．後述する計算機シミュ

3.5

RLS アルゴリズム

RLS アルゴリズムの評価関数 J は，忘却係数を λ として，レーションでの区別のため，こちらの手法を「UW 方式(11)（UW）」と表記する．なお，本節で述べる UW を用いた CLI
キャンセル手法は，第 2 章で述べた，受信部の周波数領域
等化（FDE）における UW を用いた伝送路特性推定手法とと表される．過去の信号を重みとして評価関数に組み込んは異なる．

でいる RLS アルゴリズムは，収束性が高いことが特徴としブースターにて受信信号 y(n)から取り出した UW 部分をてあげられる．更新式は以下のようになる．

UWbr(n)，ブースター内で再生した UW 真値を UWtrue(n)と(12)

すると，これらの差分が干渉成分 d(n)となる．
(8)

(13)

ここで，干渉成分 d(n)は（1）式から，親局波 s(n)に相当(14)

する UWtrue(n)を減算することで，親局波マルチパス g(n)，回り込み波 qH(n)u(n)，雑音 nAWGN(n)成分のみで構成され(15)

た UW となる．
干渉成分 d(n)は目標値として適応アルゴリズムに入力
し，再送信信号 u(n)が d(n)に近づくように適応フィルタの

ここで k(n)はゲインベクトル，P(n)は入力の相関行列の和である

タップ係数 w(n)を調整する．適応フィルタのタップの調整
は UW 部分のみで行うように制御する．

(16)

適応アルゴリズムには，提案アルゴリズムである RLS を
2 段併用した Double RLS アルゴリズムを用いる．比較アルゴリズムとして，LMS アルゴリズムと RLS アルゴリズム 6）を使用した．また，従来の UW と Data を区別せずタップ
重みを更新する GD（Data）方式と，UW 部分でのみタップ係数の更新を行う GD（UW）方式を用意した．更新式は 3.2
節従来方式 CLI キャンセラに示したものを使用した．
3.4

LMS アルゴリズムの逆行列 Φ–1(n)の推定値である．
3.6

Double RLS アルゴリズム

図 7 に RLS アルゴリズムを二つ併用した Double RLS（DRLS）アルゴリズムの構成を示す．
RLS フィルタを二つ並列で使用しており，1 段目の RLS
フィルタの出力 w1H(n)u(n)は，干渉成分 d1(n)とブースター
出力信号 u(n)を入力に用いており，通常の RLS アルゴリズ

LMS アルゴリズムによるタップ係数 w(n)の更新式は，

ムを用いた際のフィルタ出力と同じになる．

ステップサイズ µ，干渉成分 d(n)，フィルタ後出力(17)wH(n)u(n)，ブースター出力信号 u(n)を用いて，以下の式で記述される．なお，線形等化器で用いている LMS アル(18)

ゴリズムとはステップサイズや入力信号が異なる．

(19)

(9)
(10)

CLI キャンセラに用いる LMS の Step size µ は線形等化器

図 6 CLI canceller using UW for adaptive algorithm.

502 （138）

図 7 Double RLS algorithm.
映像情報メディア学会誌 Vol. 78, No. 4（2024）

Machine Translated by Google
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The BERbst is calculated using the parent station transmission bits as the true value and used

(20)

for evaluation.

(29)

In the second-stage RLS filter, the output error e1(n) in the firststage RLS filter is used as the second-stage target signal d2(n), andthe difference is compensated for by the second-stage RLS filter,

4. Computer Simulation

allowing for more accurate CLI estimation.
A computer simulation was performed using the parameters shown(twenty one)in Tables 1 to 3, assuming the use of a microwave wireless camera.
(twenty two)

Table 1. SC-FDE transmit parameters.

(twenty three)(twenty four)(twenty five)

The final output is the sum of the outputs of the two RLS filters.
It looks like this:(26)

3.7 Doppler shift
In wireless communications, when the booster or receiver itself moves, orwhen radio waves are reflected by a moving object, the frequency of the radio
waves fluctuates due to the Doppler effect. The fluctuating frequency is
calculated using the following formula. When radio waves with carrierÿfrequency fc [Hz] reacha moving object with speed v [m/s] at an angle [rad],the Doppler shift fd [Hz] is:(27)

Here, c represents the speed of light, and c ÿ 3.0×108 m/ s . For

Table 2 Booster parameters.

example, when a radio wave with fc=10 GHz is reflected from the frontÿ(=0 rad) of a moving
object approaching at v=1 m/s, fd ÿ 33 Hz.

Radio waves that have been Doppler shifted due to the Doppler
effect are recognized as a signal whose phase changes over time at
the receiving section, and the constellation rotates around the origin
at the same frequency as the Doppler shift.
To cancel the CLI, the adaptive filter of the CLI canceller needs torotate the tap coefficients w(n) on the complex plane to follow timechanges.
3.8 Evaluation
Index The evaluation of the booster’s CLI cancellation effect isbased on the modulation error ratio (MER) for F symbols calculated

Table 3 Transmission paths parameters.

from the parent station signal s(n), the linearly equalized signal l(n),and their difference eMER(n) = s(n) – l(n), as shown in Figure 3.

(28)

In addition, the booster output bit error rate (BER)
Paper ÿ Study on Coupling Interference Canceller for SC-FDE Wireless Transmission System Using High-Order Modulation(139) 503

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Rate）BERbst を，親局送信ビットを真値として計算し評価
(20)

に用いる．
(29)

2 段目の RLS フィルタでは，1 段目の RLS フィルタにおける出力誤差 e1(n)をそのまま 2 段目目標信号 d2(n)として用い，2 段目の RLS フィルタによって差分を補償することで，より正確に CLI 推定を行う．
(21)

４．計算機シミュレーション
マイクロ波帯ワイヤレスカメラの使用を前提に，表 1 〜
表 3 に示すパラメータを用いて，計算機シミュレーション(22)
表 1 SC-FDE transmit parameters.

(23)
(24)

(25)

最終的な出力は 2 段の RLS フィルタの出力の和であり，以下のようになる．
(26)

3.7

ドップラーシフト

無線通信において，ブースターや受信機自体が移動する
場合，また電波が移動体へ反射した場合，ドップラー効果
により電波の周波数が変動する．変動周波数は以下の式で計算される．速度 v[m/s]の移動体にキャリヤ周波数 fc[Hz]の電波が角度 θ [rad]で到達したときのドップラーシフト
fd[Hz]は，(27)と表される．ただし，cは光速を表し，c ≅ 3.0 × 108 m/s であ

表 2 Booster parameters.

る．例えば，fc = 10 GHzの電波がv = 1 m/sで近づいている移動体の正面（θ = 0 rad）で反射した場合，fd ≅ 33 Hzとなる．
ドップラー効果によりドップラーシフトした電波は，受信部では位相が時間的に変化している信号として認識され，コンスタレーションが原点を中心としてドップラーシ
フトと同じ周波数で回転する．ドップラーシフトを含む
CLI をキャンセルするために，CLI キャンセラの適応フィ
ルタはタップ係数 w(n)を複素平面上で時間変化に追従する
ように回転させる必要がある．
3.8

評価指標

ブースターの CLI キャンセル効果に関する評価は，図 3 に示す，親局波信号 s(n)と，線形等化後の信号 l(n)，それらの表 3 Transmission paths parameters.

差 eMER(n) = s(n) – l(n)から計算される F 個のシンボルにおける変調誤差比（MER: Modulation Error Ratio）を用いる．
(28)

また，ブースター出力ビット誤り率（BER: Bit Error論

文 □ 高次変調を行う SC-FDE 無線伝送システムのための回り込みキャンセラの検討（139） 503

Machine Translated by Google
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The occupied

0 dB, fd = 10 Hz. The CLI cancellation algorithm isbandwidth was determined based on the standard ARIB STD-B113 ). The

Comparisons were made between the GD, LMS, RLS, and DRLS of the UW method

transmission line characteristics were measured using the(labeled UW), and the GD and RLS of the data method (labeled Data). The horizontal

This was created based on the specifications of the coupling interference canceller

axis represents the number of repetitions (synonymous with the passage of time), and thefor the CU21CL terrestrial digital broadcasting relay station7 ) and the performance ofvertical axis represents MERbst , and the plots were made for the modulation method 4096QAM.

a two-stage SFN broadcast wave relay using a coupling interference canceller8 ). The

Since no significant difference was observed in the MERbst values between 256QAM

Booster input C/N, which represents the carrier-to-noise ratio, was set to 40 dB forand 4096QAM, only the results for 4096QAM are shown.

256QAM, assuming transmission under more severe conditions, since the CU21CLFirst, the steady-state MERbst is below 15 dB.

specifications7) set the standard to "equivalent C/N of the upper station signal: 45

The algorithms GD(Data), RLS(Data), and GD(UW) have many errors and are notdB." Note that for 1024QAM, which has four times the number of symbols and halves

suitable for practical use. GD(Data) has low accuracy in calculating the tap coefficients,

the distance between symbols, the noise amplitude is halved and the noise power isand GD(UW) has low accuracy in calculating the tap coefficients.
Since the number of updates for RLS(Data) is limited to the UW part, its performance

1/4, at 46 dB, and similarly for 4096QAM, it was set to 52 dB.

is even worse than that of GD(Data).

The parameters of the CLI cancellation algorithm, such as step size and RLS
forgetting factor, were set to the values that minimized BERbst when only the CLI

For (Data), the target signal for the adaptive filter must be the signal y0(n) aftercanceller was operating. The parameters of the linear equalizer were set to thesubtraction of the CLI replica, and it was found that the calculation accuracy of thevalues that minimized BERbst when the linear equalizer was operated at = 6 dB after

tap coefficients is lower than when the known signal UWtrue(n) is used to calculate

the CLI canceller
mbparameters were determined. Note that the calculation of BERbst

the target signal .

Gamma

was started after 130 blocks (approximately 40,000 bits) had elapsed, when the tap
coefficients of the CLI canceller and linear equalizer were sufficiently stable.

Next, we verify DRLS(UW), RLS(UW), and LMS(UW). The convergence speed is
DRLS(UW) ÿ RLS(UW) > LMS
It is believed that LMS (UW) had a low MERbst due to problems with steady-state
error and fluctuation tracking.

There are.

The D/U (Desired/ mb) of the two-wave model between the parent station andUndesired)(UW) has issues with tracking performance, and MERbst is lower than the highest

booster was set at 6 dB, which is approximately half theGamma

amplitude of the multipath wave g(n) compared to the desired wave (parent station

value . DRLS (UW) is able to compensate for the tracking performance of RLS (UW),and showed the best MERbst. Regarding thewave s(n)). The D/U of the CLI and the parent station wave s(n) is The CLI was set
Gammafact that no significant difference was observed between the modulation methods,

to 0 dB, assuming that the diffraction wave qH(n)u(n) has the same amplitude as theIt can be considered that there is a limit to the convergence accuracy due to thedesired wave (parent station wave s(n)).

performance of the CLI canceller.

5.1.2 CLI Doppler Frequency

5. Simulation results

Characteristics (CLI

5.1 Characteristics of the CLI Canceller

only) Next, the BERbst characteristics versus CLI Doppler frequency fd forFirst, we verified the performance of the CLI canceller when only the parent station
signal s(n) and the CLI were received at the booster receiver and only the CLI

modulation methods 256QAM to 4096QAM are shown in Figures 9 to 11. CLI, which
represents the D/U between the parent station signal s(n) and the CLI, was set to 0
Gammacanceller was operated.

dB. The cancellation algorithms compared were the data method GD (Data) and the5.1.1 Learning curve (CLI only) The
learning curve of the CLI canceller is shown in Figure 8.

fd BER for fluctuations inbst

UW method LMS (UW), RLS (UW), and DRLS (UW). RLS (Data) and GD (UW) were
Gamma

CLI

=

omitted because they had an extremely high error rate of around 0.4 to 0.5.

Figure 9 shows the fd-BERbst characteristics for 256QAM . For each algorithm, noerrors occurred in the simulation in the central area where the Doppler frequency fd
fluctuates little , and error-free transmission was possible. In terms of the relative
merits of each algorithm, DRLS (UW) showed the best characteristics, followed by
RLS (UW),The order was LMS (UW), GD (Data).
(Data), LMS (UW), RLS (UW),DRLS(UW) is winning by a large margin, and regarding DRLS(UW)The error-free interval is more than twice as wide as that of RLS (UW). This indicatesthat the second-stage tracking compensation filter of DRLS (UW) enables moreaccurate CLI tracking. Next, Fig. 10 and Fig. 11 show the fd -BERbst characteristics
Figure 8 CLI Canceller Learning Curve (4096QAM).

for 1024QAM and 4096QAM. As the modulation order increases, the fd

No multipath, no linear equalizer.

504 (140)

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0 dB, f d = 10 Hz とした．CLI キャンセルアルゴリズムは，を行った．
占有帯域幅は，標準規格 ARIB STD-B11 で規定されていUW 方式（UW と付記）の GD，LMS，RLS，DRLS および，る 値 を 用 い た 3 ）． 伝 送 路 特 性 は 日 本 通 信 機 株 式 会 社 のデータ方式（Data と付記）の GD，RLS の各方式で比較した．

CU21CL 地上デジタル放送中継局用回り込みキャンセラの横軸を繰り返し回数（時間経過と同義），縦軸を MERbst と仕様 7）および，回り込みキャンセラを用いた SFN 放送波二し，変調方式 4096QAM についてプロットした．なお，

段中継の性能 8）を参考に作成した．キャリヤ対雑音比を表

MERbst の値は 256QAM 〜 4096QAM で有意な差がみられな

す Booster input

C/N は，CU21CL の仕様 7）から「上位局信かったため，4096QAM の結果のみを示している．

号の等価 C/N ： 45 dB」を基準としていたことから，より

まず，MER bst の定常状態が 15 dB を下回っている GD

厳しい条件での伝送を想定し，256QAM で 40 dB とした．

（Data），RLS（Data），GD（UW）の各アルゴリズムに関し

なお，シンボル数が 4 倍となりシンボル間距離が 1/2 となては，エラーが多く実用には耐えられない．GD（Data）はる 1024QAM では，雑音振幅が 1/2，雑音電力が 1/4 となる

タップ係数の計算精度が低く，GD（UW）についても，GD

46 dB に，同様に 4096QAM では 52 dB とした．

（Data）の更新回数が UW 部分に制限されただけのものであStep size や RLS forgetting factor など CLI キャンセルア

るため，GD（Data）に比べてさらに性能が低くなる．RLS

ルゴリズムのパラメータは，それぞれ，CLI キャンセラの（Data）は，適応フィルタの目標信号を CLI レプリカ減算後みを動作させた状態で BERbst が最も小さくなる値とした．

の信号 y0(n)とする必要があり，目標信号の計算に既知の信線形等化器のパラメータは，CLI キャンセラのパラメータ号である UW true(n)を用いた場合と比較してタップ係数の決定後，Γmb = 6 dB で線形等化器を動作させ，BERbst が最計算精度が低くなることがわかった．

も小さくなる値とした．なお，BERbst の計算は，CLI キャ

続いて DRLS（UW），RLS（UW），LMS（UW）について

ンセラ，および，線形等化器のタップ係数が充分に安定し検証する．収束速度は，DRLS（UW）≒ RLS（UW）＞ LMS

た 130 ブロック（約 4 万ビット）経過後から計測を開始して（UW）となった．LMS（UW）は定常誤差および変動追従性いる．

の問題で，MERbst が低くなったと考えられる．また，RLS

親 局 − ブ ー ス タ ー 間 2 波 モ デ ル の D/U（ Desired /（UW）は追従性能に課題があり MERbst が最高値より低下す

Undesired）を表す Γmb は，マルチパス波 g(n)の振幅が希望

る．DRLS（UW）は，RLS（UW）の追従性能を補償できて波（親局波 s(n)）に対し約 1/2 倍となる 6 dB で伝送路モデル

おり，最も良い MERbst を示した．

を作成した．CLI と親局波 s(n)の D/U を表す ΓCLI は，回り変調方式で有意な差がみられなかったことに関しては，

込み波 qH(n)u(n)が希望波（親局波 s(n)）と同程度の振幅でCLI キャンセラの性能による収束精度の限界があると考え回り込むとして 0 dB とした．

ることができる．

５．シミュレーション結果
5.1

CLI キャンセラの特性

5.1.2 CLI ドップラー周波数 fd の変動に対する BERbst
特性（CLI のみ）
次に，変調方式 256QAM 〜 4096QAM において，CLI

まず，ブースター受信部において，親局波 s(n)と CLI のドップラー周波数 fd に対する BERbst 特性を図 9 〜図 11 に示みを受信したとし，CLI キャンセラのみを動作させた場合

す．親局波 s(n)と CLI との D/U を表す ΓCLI は 0 dB とした．

の CLI キャンセラの性能を検証した．

キャンセルアルゴリズムはデータ方式の GD（Data），UW

5.1.1

学習曲線（CLI のみ）

CLI キャンセラの学習曲線を図 8 に示す．CLI は，ΓCLI =方式の LMS（UW），RLS（UW），および，DRLS（UW）で比較した．RLS（Data），GD（UW）はエラーレートが 0.4 〜
0.5 程度とエラーが非常に多い状態だったため省略する．
図 9 に 256QAM における fd – BERbst 特性を示す．各アルゴリズムにおいて，ドップラー周波数 fd の変動が少ない中央部分は，シミュレーションでエラーが生じず，エラーフ
リーな伝送が行えている．各アルゴリズムの優劣について
は，DRLS（UW）が最も良い特性を示し，次いで RLS（UW），
LMS（UW），GD（Data）の順になった．特に従来方式の GD（Data）に対し，UW 方式の LMS（UW），RLS（UW），
DRLS（UW）は大きく勝っており，DRLS（UW）に関しては
RLS（UW）に対し 2 倍以上エラーフリーな区間が広い．これは，DRLS（UW）の 2 段目の追従補償フィルタにより，より正確に CLI 追従が行えるようになったことを示している．

図 8 CLI Canceller Learning Curve（4096QAM）.

続いて，図 10，図 11 に 1024QAM，4096QAM におけるno multipath, no linear equalizer.

f d – BERbst 特性を示す．変調次数が大きくなるにつれて f d

504 （140）

映像情報メディア学会誌 Vol. 78, No. 4（2024）

Machine Translated by Google
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The above considerations will be discussed in Section 5.2.2 Changes in BERbst with Number of Taps.

5.2 Linear equalizer + CLI canceller Next, we
assumed that the booster receiver received the parent station signal s(n) and
CLI as well as the parent station signal multipath g(n), and verified the signal
convergence performance and BERbst characteristics against the Doppler shift fd
when the linear equalizer was operated in addition to the CLI canceller.

5.2.1 Learning Curve (CLI + Multipath)
The learning curves when both the CLI canceller and the linear equalizer are
operated are shown in Figures 12 to 14. The modulation method is 256QAM~ = 0

Figure 9 CLI Doppler fd vs. BERbst (1024QAM).
No multipath, no linear equalizer.

4096QAM, CLI = 6 dB.

Gamma

CLI

dB, fd = 10 Hz, and multipath is

Gamma

mb

First, in the learning curve for 256QAM in Fig. 12, the convergence of the tap
coefficients to the optimal value increases in two stages.
In the first rise up to around 0.1×10-6, CLI is eliminated mainly by the operation of
the CLI canceller, and the convergence is rapid. After that , when the number oflearning times is between 0.1×10-6 and 0.5×10-6, the parameters for eliminating
the remaining multipath waves are linearly equalized.
However, the linear equalizer is learning the domain.
The step size is small, which slows down the convergence of multipath and makes
the overall learning characteristics gentler. The second steep rise in the number oflearning iterations from 0.5×10–6 to 0.6×10–6 indicates that the linear equalizer is
now able to equalize multipath to a certain extent.
It is believed that this allows the CLI canceller to create an accurate replica of the
CLI, which in turn allows the MERbst value to converge further.

Figure 10 CLI Doppler fd vs. BERbst (1024QAM).
No multipath, no linear equalizer.

In the learning curves for 1024QAM and 4096QAM shown in Figures 13 and 14,
the number of learning iterations rises rapidly to around 0.1 × 10–6 , after which
the slope becomes gentler. This is thought to be because, initially, rapid
convergence occurs due to the operation of the linear equalizer and CLI canceller,
but in the subsequent learning of the linear equalizer, learning does not progresssmoothly and takes time due to factors such as the inability to reduce demodulation
errors in the re-modulator section, resulting in a drop in learning efficiency.

Figure 11 CLI Doppler fd vs. BERbst (4096QAM).
No multipath, no linear equalizer.

It can be seen that the degradation of the BERbst value for fd becomes greaterand the error-free interval becomes narrower. This is because there is an issue
with the CLI cancellation performance, and the tracking and convergence
performance for Doppler frequency fluctuations is insufficient, resulting indemodulation errors in the demodulation, decision and

Figure 12 CLI Canceller Learning Curve (256QAM).

remodulation blocks. Also, there is a difference in BERbst depending on whether fd is positive or negative.

with multipath, with linear equalizer.

(141) 505
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Paper ÿ Study on Coupling Interference Canceller for SC-FDE Wireless Transmission System Using High-Order Modulation

ての考察は 5.2.2 節タップ数による BERbst の変化で述べる．
5.2

線形等化器 + CLI キャンセラ

次にブースター受信部において，親局波 s(n)と CLI に加えて，親局波マルチパス g(n)を受信したとし，CLI キャン
セラだけでなく線形等化器も動作させた際の信号収束性能
およびドップラーシフト fd に対する BERbst 特性について検
証した．
5.2.1

学習曲線（CLI + マルチパス）

CLI キャンセラと線形等化器の両方を動作させた際の学習曲線を図 12 〜図 14 に示す．変調方式は，256QAM 〜
図 9 CLI Doppler fd vs. BERbst（1024QAM）.
no multipath, no linear equalizer.

4096QAM，CLI は ΓCLI = 0 dB, fd = 10 Hz，マルチパスは

Γmb = 6 dB とした．
まず，図 12 の 256QAM における学習曲線では，タップ係数の最適値への収束が 2 段階で上昇している．学習回数が 0.1 × 10–6 近辺までの 1 回目の上昇では，主に CLI キャン
セラの動作により CLI が除去され急速に収束している．その後の，学習回数が 0.1 × 10–6 〜 0.5 × 10–6 にかけては，残りのマルチパス波を除去するためのパラメータを線形等化器が学習している領域となっている．ただ，線形等化器のStep size は小さくしている影響もあり，マルチパスの収束が遅くなり全体の学習特性も緩やかになっている．学習回
数が 0.5 × 10 –6 〜 0.6 × 10 –6 の 2 回目の急峻な上昇は，線形等化器がある程度マルチパスを等化できるようになったことで，CLI キャンセラにより CLI の正確なレプリカを作成でき， MER bst の値をさらに収束させることができるよう

図 10

CLI Doppler fd vs. BERbst（1024QAM）.
no multipath, no linear equalizer.

になったと考えられる．
図 13，図 14 に示す 1024QAM，4096QAM における学習曲線では，学習回数が 0.1 × 10–6 近辺まで急速に上昇したの
ち，傾きが緩やかになっている．これは，初めに線形等化器と CLI キャンセラの動作により急速な収束が生じるが，その後の線形等化器の学習において，Re-modulator 部での復調エラーが減らないなどの影響により学習が中々進まずに時間がかかり，学習効率が落ちていると考えられる．

図 11

CLI Doppler fd vs. BERbst（4096QAM）.
no multipath, no linear equalizer.

に対する BERbst の値の劣化が大きくなり，エラーフリーの区間も狭くなることがわかる．これは CLI キャンセル性能に課題があり，ドップラー周波数変動への追従性能と収束
性能が足りておらず，復調・判定・再変調ブロックで復調
エラーが生じているためであると言える．
また fd の正負によって BERbst に差があるが，これについ論

文 □ 高次変調を行う SC-FDE 無線伝送システムのための回り込みキャンセラの検討

図 12

CLI Canceller Learning Curve（256QAM）.
with multipath, with linear equalizer.

（141） 505

Machine Translated by Google
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Figure 15 CLI Doppler fd vs. BERbst (256QAM).
with multipath, with linear equalizer.

Figure 13 CLI Canceller Learning Curve (1024QAM).
with multipath, with linear equalizer.

Figure 16 CLI Canceller Output Phase Difference (256QAM).

Figure 14 CLI Canceller Learning Curve (4096QAM).
with multipath, with linear equalizer.

5.2.2 CLI Doppler Frequency

fd BER characteristics forbstvariations in (CLI + multipath)
Next, Fig. 15 to Fig. 17 show the modulation methods 256QAM to 4096QAM.
fd –BERbst when the Doppler frequency fd of the CLI is changed
The characteristics of CLI and multipath are as follows:
dB,

Gamma

CLI

=0

mb = 6 dB. At a transmission frequency of fc = 10 GHz,

Gamma

the Doppler frequency fd = 20 Hz is when the speed of the moving object is v ÿThis corresponds to 0.6 m/s ÿ 2.16 km/h.
First, in Figure 15, which shows the fd –BERbst characteristics of 256QAM,
The error-free frequency range is –9 to +7 Hz for DRLS, which is the best.
This compares favorably with the RLS, which has the next best characteristics in the range of –3 to +3 Hz.

Figure 17 CLI Doppler fd vs. BERbst (1024QAM).
with multipath, with linear equalizer.

The CLI canceller in Section 5.1.2 is used.
When compared with the fd –BERbst characteristics when the
Therefore, this difference is linear, etc.

This is due to the differences that exist.
Next , in Figure 17, which shows the fd-BERbst characteristics of 1024QAM ,This can be said to be due to the degradation caused by the multipath equalization performance of the equalizer.

The CLI cancellation algorithm also works well when the BERbst is 256QAM.

The BERbst is asymmetric between positive and negative fd.

The deterioration is compared with the RLS (UW) case near fd=0 Hz.

This is because the learning process of the tap changes depending on whether it is positive or negative, and as shown in Figure 16,

The superiority of DRLS (UW) is reversed, but this is due to the difference in the modulation method.

As shown in the figure, the phase tracking of the CLI canceller output is positive and negative.

This is because the optimal step size µm of the linear equalizer changes with506 (142)

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図 15

CLI Doppler fd vs. BERbst（256QAM）.

with multipath, with linear equalizer.

図 13

CLI Canceller Learning Curve（1024QAM）.
with multipath, with linear equalizer.

図 16

図 14

CLI Canceller Output Phase Difference（256QAM）.

CLI Canceller Learning Curve（4096QAM）.
with multipath, with linear equalizer.

5.2.2 CLI ドップラー周波数 fd の変動に対する BERbst
特性（CLI + マルチパス）
続いて図15〜図17に，変調方式256QAM〜4096QAMにおいて CLI のドップラー周波数 fd を変動させた際の fd – BERbst
特性を示す．CLI およびマルチパスは，それぞれ，ΓCLI = 0
dB, Γmb = 6 dB とした．なお，送信周波数 fc = 10 GHz におけるドップラー周波数 f d = 20 Hz は，移動体の速度が v ≅0.6 m/s ≅ 2.16 km/h に相当する．
まず，256QAM の fd – BERbst 特性を示した図 15 において，
エラーフリーの周波数範囲は DRLS が –9 〜 +7 Hz と最も優れており，次に良好な特性を示す RLS の – 3 〜 +3 Hz に対

図 17

CLI Doppler fd vs. BERbst（1024QAM）.

with multipath, with linear equalizer.

し 2 倍以上広くなっている．5.1.2 節の CLI キャンセラのみを動作させた際の fd – BERbst 特性（図 9）と比較すると，全差が生じていることに起因している．

体的に BERbst が劣化している．そのため，この差は線形等

次に 1024QAM の fd – BERbst 特性を示した図 17 では，ど化器のマルチパス等化性能による劣化だと言える．また，の CLI キャンセルアルゴリズムも BER bst が 256QAM の場

正の fd と負の fd で左右非対称の BERbst となっているが，こ合と比較して劣化している．fd = 0 Hz 付近で RLS（UW）とれはタップの学習過程が正負で変化することや，図 16 にDRLS（UW）の優劣が逆転しているが，これは各変調方式示すに示すように CLI キャンセラ出力の位相追従に正負でで線形等化器の最適な Step size µm が変化するためである

506 （142）

映像情報メディア学会誌 Vol. 78, No. 4（2024）

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Figure 19 RLS Taps vs. BERbst (256QAM).
Figure 18 CLI Doppler fd vs. BERbst (4096QAM).

in each Linear Equalizer tap.

with multipath, with linear equalizer.

it is conceivable that.

In addition, in Figure 18, which shows the fd – BERbst characteristics for
4096QAM , the BERbst is significantly worse for all algorithms than for 256QAM
and 1024QAM. This is because, as shown in Figures 11 and 14, the convergenceaccuracy of the CLI canceller and the multipath equalization processing capabilityof the linear equalizer are bottlenecks.

5.2.3 Changes due to tap number
The number of taps in the linear equalizer was changed for each of the RLS(UW) and DRLS (UW) algorithms, and the relationship between the number of taps
D in the CLI canceller and BERbst was examined. The results are shown in Figures
19 to 24.

Figure 20. RLS Taps vs. BERbst (1024QAM).
in each Linear Equalizer tap.

For 4096QAM, the number of taps of the linear equalizer (LEq taps) is set to 5 to
35 taps, and the number of taps of the CLI canceller D is set to 1 to 30 taps.
The BERbst is plotted for each of the following: = 0 dB, fd = 10 Hz, multipath wave
CLI

Gamma

CLI

Gamma

mb

= 6 dBIt was decided.

First, when the CLI canceller tap number D is one (the horizontal axis value is1) , both the RLS and DRLS algorithms achieve 256QAM to
Cancellation performance is not fully demonstrated for all modulation formats of
4096QAM. This is because the output signal u(n) is buffered by the number oftaps D and used in the cancellation calculation. When the CLI delay in thesimulation conditions is 2 samples, the number of taps D must be two or more inorder for CLI to be cancelled. In this way, the length of CLI delay that can becancelled is equal to the number of taps D of the CLI canceller.

In Figures 19 to 21, which show the results of the RLS algorithm, the BERbst

Figure 21 RLS Taps vs. BERbst (4096QAM).

characteristic is flat when the CLI canceller tap count D is 2 to 30 , and it can bein each Linear Equalizer tap.

seen that the BERbst value depends only on the tap count of the linear equalizer.
The more taps the linear equalizer has, the higher the BERbst improves, which is

There is no significant difference between the number of taps of the filter being 30 and 35.

because the linear equalizer

The BERbst value is saturated at around 3×10–3. For

This is because the equalization performance improves as the number of taps
increases.
In the Taps-BERbst characteristic of 256QAM shown in Figure 19 , the linear

1024QAM and 4096QAM shown in Figures 20 and 21, the degradation of the
BERbst value increases as the modulation order increases. This is similar to theresults shown in the fd-BERbst characteristics in Figures 16 and 17 , and is due to the CLI.

(143) 507
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Paper ÿ Study on Coupling Interference Canceller for SC-FDE Wireless Transmission System Using High-Order Modulation

図 19

図 18 CLI Doppler fd vs. BERbst（4096QAM）.

RLS Taps vs. BERbst（256QAM）.
in each Linear Equalizer tap.

with multipath, with linear equalizer.

と考えられる．
また，4096QAM の f d – BER bst 特性を示した図 18 では，すべてのアルゴリズムで BERbst が 256QAM，1024QAM の場合に比べ大きく劣化している．これは，図 11 や図 14 に示すように，CLI キャンセラの収束精度や線形等化器におけるマルチパス等化処理能力がボトルネックになっているためである．
5.2.3

タップ数によるの変化

RLS（UW），DRLS（UW）それぞれのアルゴリズムにおいて，線形等化器のタップ数を変えて，CLI キャンセラのタップ数 D に対する BERbst の関係について検討した．その結果を図 19 〜図 24 に示す．変調方式 256QAM 〜

図 20

4096QAM において，線形等化器のタップ数（LEq taps）を

RLS Taps vs. BERbst（1024QAM）.
in each Linear Equalizer tap.

5 〜 35 タップ，CLI キャンセラのタップ数 D を 1 〜 30 タップまでそれぞれ変化させた際の BER bst をプロットした．
CLI は ΓCLI = 0 dB, fd = 10 Hz，マルチパス波は Γmb = 6 dBとした．
まず，CLI キャンセラのタップ数 D が一つ（横軸の値が 1）の場合，RLS，DRLS 両アルゴリズムとも 256QAM 〜
4096QAM のすべての変調方式においてキャンセル性能がまったく発揮できていない．これは，出力信号 u(n)をタッ
プ数 D 分バッファしキャンセル計算に用いており，シミュ
レーション条件の CLI 遅延が 2 sample の場合においては，
タップ数 D が二つ以上ないと CLI をキャンセルすることができないためである．このように，キャンセル可能な CLI
遅延の長さは，CLI キャンセラのタップ数 D と等しくなる．
RLS アルゴリズムの結果を示した図 19 〜図 21 では，CLI

図 21

RLS Taps vs. BERbst（4096QAM）.
in each Linear Equalizer tap.

キャンセラのタップ数 D が 2 〜 30 において BERbst 特性がフラットになっており，BERbst 値は線形等化器のタップ数のみに依存していることがわかる．線形等化器のタップ数は化器のタップ数が 30 の場合と 35 の場合で大きな差がなく，多いほど BER bst が向上しており，これは線形等化器がBERbst 値が 3 × 10–3 付近で飽和している．

FFE（Feed Forward Equalizer）であり，タップ数の増加に伴い等化性能が向上することに起因する．
図 19 に示す 256QAM の Taps – BERbst 特性では，線形等論

図 20，図 21 に示す 1024QAM，4096QAM では，高次の変調になるにつれ BERbst の値の劣化は大きくなる．これは，
図 16，図 17 の fd – BERbst 特性に示した結果と同様に，CLI

文 □ 高次変調を行う SC-FDE 無線伝送システムのための回り込みキャンセラの検討（143） 507

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This is due to the convergence accuracy of the canceller and the performance of theThis is because the tap coefficients of the taps that do not actually need to operate (inmultipath equalization process in the linear equalizer, and it is believed that improving

this case, all taps except the second tap) are also adjusted, which becomes noise andthese will also improve the BERbst value.

makes it impossible to generate a CLI replica correctly. Therefore, the best BERbst

In Figures 22 to 24, which show the results of the DRLS algorithm,

value is obtained when there are only two taps, the minimum number required for CLI

The values are generally improved compared to the RLS algorithm. However, thecancellation.

values deteriorate as the number of CLI canceller taps increases.

The reason why this phenomenon does not occur with the RLS algorithm is that theconvergence performance of the RLS algorithm itself is poor, rather than the degradationof the BERbst value due to the operation of tap coefficients that do not need to be
operated.
In summary, the linear equalizer is an FFE, so themore taps it has, the better its performance. The CLI canceller is the main

The value is improved by including the CLI delay time in the number of taps.
It turns out there is a connection.

3. Conclusion
We investigated a CLI canceller for a SC-FDE wireless transmission system using
Fig. 22 DRLS Taps. BERbst (256QAM). in eachLinear Equalizer tap.

high-order modulation for professional wireless cameras using microwave bands. We
proposed a CLI canceller that uses only UW for learning in the SC-FDE system, and a
new Double RLS algorithm that uses two stages of RLS. In a transmission simulation
using 256QAM, we confirmed that Double RLS has more than twice the CLI Doppler
tolerance of RLS.

In the future, we will continue to improve signal processing accuracy and multipath
equalization and loop interference cancellation functions in order to realize a booster
with a CLI canceller that can also support 1024QAM and 4096QAM.

This research was funded by the Broadcasting Culture Foundation in fiscal 2019 andThis project was carried out with the support of grants in fiscal year 2022 and 2023.

[References]

Figure 23. DRLS Taps vs. BERbst (1024QAM).

1) Y. Matsusaki, F. Yamagishi, A. Yamasato, T. Nakagawa, S. Okabe and N. Iai:
"Development of UHDTV Wireless Camera Transmitter using Millimeter-Wave Band",in each Linear Equalizer tap.

IEEE RWS 2020, pp.24-26 (2020)
2) T. Kimura, H. Ogawa, T. Suzuki and H. Hamazumi: "Study on Single Frequency
Downlink with Coupling Loop Interference Canceler for Professional SC-FDE WirelessCamera using Millimeter-wave Band", ITE Trans. on MTA 11, 1, pp.13-25 (Jan. 2023)

3) Association of Radio Industries and Businesses: “Portable microwave digital wireless transmission
system for transmission of television broadcasting program material”, ARIB STD-B11 version 2.2(Nov. 2005) 4) Association of Radio Industries and Businesses: “Portable microwave OFDM digital wirelesstransmission system for transmission of ultra-high definition television broadcasting program material”,
ARIB STD-B71 version 1.0 (Jan. 2018)5) DC Chu: "Polyphase Codes with Good Periodic Correlation properties: "IEEE Trans,
Info. Theory, pp.531-532 (July1972)
6) S. Haykin: "Adaptive filter theory", 4th Edition, Prentice Hall, New
Jersey (2002) 7)Japan Communication Equipment Co., Ltd.: “ISDB-T terrestrial digital broadcasting relay
station equipment CU21CL: Loop canceller (H=50mm)”, Japan Communication
Equipment Co., Ltd., https:// nitsuki.com/products/broadcast/sate/single/CU21CL.html(accessed 2023.10.16)
8) Hiroyuki Hamazumi, Koichiro Imamura, Kohei Kanbara, Akihiro Yasuda, Kazuhiko Shibuya, Hiroo

Figure 24 DRLS Taps vs. BERbst (4096QAM).
in each Linear Equalizer tap.

508 (144)

Arata, Dan Saeki, Kazuyuki Yoshida, Makoto Sasaki, “Performance of SFN Broadcast Wave
Two-Stage Relay Using a Loop Canceller”, ITE Technical Report, 26, 32, pp.13-16 (May. 2002)

Journal of the Institute of Image Information and Television Engineers Vol. 78, No. 4 (2024)【Copyrights to Machine-Translated Content】
The copyright of the original papers published on this website belongs to the Institute of Image Information and Television Engineers.
Unauthorized use of original papers or translated papers is prohibited.
For details, please refer to the copyright regulations of the Institute of Image Information and Television Engineers.

キャンセラの収束精度や線形等化器におけるマルチパス等化していく．これは，本来動作する必要のないタップ（今

化処理の性能に起因し，これらを改善することで BERbst 値

回は 2 タップ目以外すべてのタップ）のタップ係数にも調も改善すると考えられる．

整が加わり，それが雑音となり CLI レプリカを正しく生成

DRLS アルゴリズムの結果を示した図 22 〜図 24 では，できなくなるからである．そのため CLI キャンセルに最低RLS アルゴリズムと比較して全体的に値が向上している．

限必要なタップ二つの場合が最も良い BER bst 値を示す．

しかし，CLI キャンセラのタップ数の増加とともに値が劣RLS アルゴリズムでこの現象が見られない理由は，本来動作する必要のないタップ係数の動作による BERbst 値の劣化よりも，RLS アルゴリズム自体の収束性能が悪く，タップ係数の影響よりも RLS アルゴリズムの影響によりもともと
BERbst 値が劣化していると言える．
まとめると，線形等化器は FFE であることもありタップ
数は多いほど性能がよくなり，CLI キャンセラは主要な
CLI の遅延時間を包含するタップ数とすることで値向上につながることがわかった．

３．む

す

び

マイクロ波帯を用いた業務用ワイヤレスカメラに向け
図 22

DRLS Taps. BERbst（256QAM）.
in each Linear Equalizer tap.

た，高次変調を行う SC-FDE 無線伝送システムのための回り込み（CLI）キャンセラの検討を行った．SC-FDE 方式において，UW のみを使用し学習を行う CLI キャンセラを提案するとともに，RLS を 2 段併用した新たな Double RLS
アルゴリズムを提案した．256QAM を用いた伝送シミュ
レーションにおいては，Double RLS が RLS に対して 2 倍
以上の CLI ドップラー耐性を持つことを確認した．
今後は 1024QAM，4096QAM にも対応できる CLI キャン
セラ付きブースターの実現に向けて，信号処理精度との向上とマルチパス等化および回り込みキャンセル機能の改善を行っていく．
本研究は，公益財団法人放送文化基金から 2019 年度および 2022 年度の助成を受けて実施した．

〔文
図 23

DRLS Taps vs. BERbst（1024QAM）.
in each Linear Equalizer tap.

図 24

DRLS Taps vs. BERbst（4096QAM）.
in each Linear Equalizer tap.

508 （144）

献〕

1）Y. Matsusaki, F. Yamagishi, A. Yamasato, T. Nakagawa, S. Okabe
and N. Iai: "Development of UHDTV Wireless Camera Transmitterusing Millimeter-Wave Band", IEEE RWS 2020, pp.24-26 (2020)2）T. Kimura, H. Ogawa, T. Suzuki and H. Hamazumi: "Study on Single
Frequency Downlink with Coupling Loop Interference Canceler forProfessional SC-FDE Wireless Camera using Millimeter-wave Band",ITE Trans. on MTA 11, 1, pp.13-25 (Jan. 2023)3）電波産業会： テレビジョン放送番組素材伝送用可搬形マイクロ波帯
デジタル無線伝送システム ，ARIB STD-B11 version 2.2（Nov. 2005）
4）電波産業会： 超高精細度テレビジョン放送番組素材伝送用可搬形マ
イクロ波帯 OFDM 方式デジタル無線伝送システム ，ARIB STD-B71
version 1.0（Jan. 2018）5）D.C. Chu: "Polyphase Codes with Good Periodic Correlation
properties: "IEEE Trans, Info. Theory, pp.531-532 (July1972)6）S. Haykin: "Adaptive filter theory", 4th Edition, Prentice Hall, New
Jersey（2002）
7）日本通信機株式会社： ISDB-T 地上デジタル放送中継局用機器
CU21CL:回り込みキャンセラー（H=50mm），日本通信機，https://nitsuki.com/products/broadcast/sate/single/CU21CL.html（2023.10.16 参照）
8）濱住啓之，今村浩一郎，神原浩平，安田明広，渋谷一彦，阿良田洋雄，佐伯暖，吉田和幸，佐々木誠： 回り込みキャンセラを用いた SFN
放送波二段中継の性能 ，映情学技報，26，32，p.13-16（May. 2002）

映像情報メディア学会誌 Vol. 78, No. 4（2024）

Machine Translated by Google
This is a machine-translated version of the original paper in English. The next page is the original paper, which is expanded alternately with the translated version.
Because this is a machine translation, it may contain typographical errors, mistranslations, or parts of the paper that have not been reflected in the translation.

Tomoyuki Suzuki
Tomoyuki Suzuki graduated from the Department of Electrical Engineering at Tokyo National College of Technology in 2022. In thesame year, he enrolled in the school's specialized course in electrical and electronic engineering. In 2024, he completed his studies inthe specialized course in electrical and electronic engineering. He currently works for SoftBank Corp.

Masato Yokoi
Masato Yokoi graduated from the Department of Electrical Engineering at the Tokyo National College of Technology in 2023. He iscurrently enrolled in the Department of Mathematics at the School of Science at the Tokyo Institute of Technology.

Hotaka Ogawa
Hodaka Ogawa Graduated from the Department of Electrical Engineering at Tokyo National College of Technology in 2022. Enrolled in
the school's specialized course in electrical and electronic engineering in the same year. Graduated from the school's specialized
course in electrical and electronic engineering in 2024. Currently working for ST Microelectronics Corporation.

Hiroyuki Hamazumi
Hiroyuki Hamazumi Graduated from the Department of Electrical Engineering at Fukui National College of Technology in 1982. He
joined NHK in the same year. After working at Tsu Broadcasting Station, he has worked at the NHK Science and Technology ResearchLaboratories since 1987. Since then, he has been engaged in research into highly reliable digital wireless transmission, such astransmission and reception technology for terrestrial digital broadcasting and wireless transmission of program material. Since 2018,he has been a professor in the Department of Electrical Engineering at Tokyo National College of Technology. Since 2023, he has beena professor in the Department of Electrical and Electronic Engineering at Fukui National College of Technology. Doctor of Engineering.
Fellow certified member of our association.

Tomohiko Kimura
Tomohiko Kimura completed his studies in electronic and physical engineering at the Graduate School of Pure and Applied Sciences,
University of Tsukuba in 2005. In 2006, he became a research assistant and assistant professor in the Department of ElectricalEngineering at the Tokyo National College of Technology. In 2013, he became an associate professor in the Department of ElectricalEngineering at the same college. PhD (Engineering). Regular member.

(145) 509
【Copyrights to Machine-Translated Content】
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Paper ÿ Study on Coupling Interference Canceller for SC-FDE Wireless Transmission System Using High-Order Modulation

す ず き

ともゆき

鈴木

智之

2022 年，東京工業高等専門学校電気工学科卒業．同年，同校専攻科電気電子工学専攻入学．
2024 年，同校専攻科電気電子工学専攻修了．現在，ソフトバンク（株）に勤務．

よ こ い

ま さ と

横井

将人

2023 年，東京工業高等専門学校電気工学科卒業．現在，東京工業大学理学院数学系在学中．

お が わ

ほ た か

小川

穂高

2022 年，東京工業高等専門学校電気工学科卒業．同年，同校専攻科電気電子工学専攻入学．
2024 年，同校専攻科電気電子工学専攻修了．現在，ST マ
イクロエレクトロニクス（株）に勤務．

はまずみ

ひろゆき

濱住

啓之

1982 年，福井工業高等専門学校電気工学科卒業．同年，NHK 入局．津放送局を経て，1987 年より，同放送技術研究所に勤務．以来，地上デジタル放送の送受信技術や番組素材無線伝送など，高信頼ディジタ
ル無線伝送の研究に従事．2018 年より，東京工業高等専
門学校電気工学科教授．2023 年より，福井工業高等専門
学校電気電子工学科教授．工学博士．当会フェロー認定
会員．
き む ら

ともひこ

木村

知彦

2005 年，筑波大学大学院数理物質研究科電子・物理工学専攻修了．2006 年，東京工業高等専門
学校電気工学科助手，助教を経て，2013 年より，同校電気工学科准教授．博士（工学）．正会員．

論

文 □ 高次変調を行う SC-FDE 無線伝送システムのための回り込みキャンセラの検討（145） 509