QDXQDX: QRP Labs Digital Xcvr (transceiver)
Assembly, design and performance manual--- PCB Rev 5 ---QDX assembly Rev 1.18d

1Contents
Contents.......................................................................................................................................................................... 2
1. Introduction.................................................................................................................................................................3
2. Assembly.....................................................................................................................................................................5
2.1 General guidelines.................................................................................................................................................5
2.2 Parts list.................................................................................................................................................................9
2.3 Inventory parts....................................................................................................................................................13
2.4 Install all the ceramic capacitors..........................................................................................................................14
2.5 Install 1N4007 diodes..........................................................................................................................................16
2.6 Install 47uH inductors..........................................................................................................................................17
2.7 Install PA transistors............................................................................................................................................18
2.8 Assemble and install transformer T1...................................................................................................................19
2.9 Prepare and install tapped inductor L12..............................................................................................................24
2.10 Wind and install L14..........................................................................................................................................30
2.11 Install Low Pass Filter toroids............................................................................................................................31
2.12 Wind and install trifilar toroid T2.......................................................................................................................33
2.13 Install the 3mm red status LED..........................................................................................................................36
2.14 Install jumper wire.............................................................................................................................................37
2.15 Install connectors..............................................................................................................................................38
2.16 Optional enclosure............................................................................................................................................39
3 Design......................................................................................................................................................................... 41
3.1 Summary.............................................................................................................................................................41
3.2 Block Diagram......................................................................................................................................................44
3.3 Synthesized local oscillator..................................................................................................................................45
3.4 Solid state transmit/receive switch.....................................................................................................................49
3.5 Switched receiver band pass filter.......................................................................................................................50
3.6 Double-balanced Quadrature Sampling Detector................................................................................................51
3.7 Low noise balanced differential pre-amplifiers....................................................................................................52
3.8 Analog to Digital Converter.................................................................................................................................53
3.9 Embedded Software Defined Radio.....................................................................................................................54
3.10 Embedded 24-bit 48ksps stereo USB Sound card..............................................................................................57
3.11 CAT control serial interface...............................................................................................................................58
3.12 Audio frequency analysis...................................................................................................................................58
3.13 Class-D Push-pull Power amplifier.....................................................................................................................68
3.14 Switched output Low Pass Filters......................................................................................................................69
3.15 Configurable PTT (Push To Talk) output............................................................................................................87
3.16 Voltage regulation and supply decoupling.........................................................................................................88
4 Performance measurements......................................................................................................................................89
4.1 Receive current consumption..............................................................................................................................89
4.2 Transmit current consumption............................................................................................................................89
4.3 Power output vs supply voltage..........................................................................................................................90
4.4 Output harmonic content....................................................................................................................................91
4.5 Unwanted sideband suppression........................................................................................................................92
5. Resources.................................................................................................................................................................. 93
6. Document Revision History.......................................................................................................................................93

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1.

Introduction

The QDX is a high performance, five-band 5W Digital modes transceiver with CAT control andbuilt-in USB sound card. QRP Labs presents QDX, a digital transceiver with a ratio of performance
to price not available until now.
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Versions: 80, 60, 40, 30 and 20m OR 20, 17, 15, 12, 11 and 10m; 5 W from 9 – 10V supply
Clean single signal output (zero residual carrier, zero unwanted sideband)
Solid state PIN-diode switched Low Pass Filters and solid state Band Pass Filters
Solid state transmit/receive switching
High performance embedded-SDR SSB receiver using 112dB 24-bit stereo ADC chip
Built-in USB sound card: 48ksps 24-bit stereo
Built in USB Virtual COM port serial for CAT control
Si5351A Synthesized local oscillator with better than 0.001Hz resolution and high precision
25MHz TCXO reference as standard
Built-in signal generator
Built-in suite of configuration and analysis tools
Lifetime free firmware upgrades with QRP Labs Firmware Update (QFU) bootloader foreasy firmware update on any OS with no extra software, or drivers, or programming
hardwareAll SMD components pre-installed by factory, only through-hole component soldering by the
kit constructor
Receive current: 150mA; Transmit current 1.0 – 1.1A (9V supply, 5W output)
Only four connectors: USB (audio and serial for CAT), Power, PTT (external amp) and RF
Optional smart aluminium extruded enclosure measuring just 89 x 63 x 25mm

No test equipment is required to build, align and operate this digi modes transceiver. There are noalignment tasks.
We hope you enjoy building and operating this kit! Please read this manual carefully, and follow
the instructions step by step in the recommended order. Later in the manual the circuit design isdescribed in detail and we recommend reading and understanding this section too, to get the
maximum enjoyment and education from your new radio.
Typical performance measurements are shown in the measurements section.
The operating manual is a separate document and will get you started with QDX and your
WSJT-X or other digi modes software in minutes.

PLEASE READ THE BASIC ASSEMBLY AND USE INSTRUCTIONS IN
THIS MANUAL VERY CAREFULLY BEFORE APPLYING POWER TO
THE BOARD!
Additional connections:
The Rev 5 PCB has additional pin header pads on the top right corner of the PCB (as pictured)
which provide access to the VGA monitor and PS/2 keyboard ports, and serial port #2. The PTT
connection can be re-purposed as serial port #3. Details of these functions are in the operating
manual. They’re just easier to access now on the Rev 5 PCB than on earlier PCB revisions.

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IMPORTANT!
QDX can be built for 9V or 12V operation! You need to decide NOW!
A 9V QDX produces 5 W power output from a supply voltage of 9V or a little over. At 12V, a QDX
built for 9V could be producing 8 W power output which is likely to cause over-heating andperhaps failure of the BS170 final transistors. Do not power a QDX with a higher voltage than you
built it for.
Operation of QDX at more than 6W power output is NOT RECOMMENDED.

If you wish to operate using a 12 V supply, you may use a two turn secondary winding on the
output transformer T1, so a 3:2 ratio instead of the 3:3 turns ratio documented in this manual.
Remember this when you come to the assembly step for preparing and installing the output
transformer T1. The “primary” is still 3 turns, with a tap half way at 1.5 turns. The secondary (notap) will now be only two turns.
The chart below shows the measured power output vs supply voltage for the standard 3:3 winding(Red line); at 12 V supply the output power of around 8 W is too high and likely to cause overheating or failure of the power amplifier transistors. If you wish to use a supply of 12 - 13 V the 3:2
winding style is more suitable and will produce 4 – 5 W output for 12 – 13 V supply. The graphshows 40m but other bands are very similar.
Operation at more than 12V will reduce the protection safety margin available in the event of very
long key-downs, hot environment, antenna mismatch, cable faults etc. I don’t recommend morethan 12V. It isn’t worth pushing more just for a fraction of a dB. It is acceptable to use diodes inseries on the power line, each one will drop the voltage 0.6V or so. Beware that a 12V nominal
batter will potentially be several volts higher when fully charged.

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2.

Assembly

2.1 General guidelines
Assembly of this kit is quite straightforward, most components are SMD and have already been
pre-assembled by the PCB factory. The usual kit-building recommendations apply: work in a welllit area, with peace and quiet to concentrate. Some of the other semiconductors in the kit are
sensitive to static discharge. Therefore, observe Electrostatic discharge (ESD) precautions.
And I say it again: FOLLOW THE INSTRUCTIONS!!
Don’t try to be a hero and do it without instructions!
A jeweler's loupe is really useful for inspecting small
components and soldered joints. You’ll need a finetipped soldering iron too. It is good to get into the habit ofinspecting every joint with the magnifying glass or
jeweler's loupe (like this one I use), right after soldering.
This way you can easily identify any dry joints or solder
bridges, before they become a problem later on when
you are trying to test the project.
You could also take photos with a mobile phone, and use
the phone’s zoom features to view the board in detail.
Triple check every component value and location BEFORE soldering the component!
It is easy to put component leads into the wrong holes, so check, check and check again! It isdifficult to de-solder and replace components, so it is much better to get them correctly installed
the first time. In the event of a mistake, it is always best to detect and correct any errors as earlyas possible (immediately after soldering the incorrect component). Again, a reminder: removing acomponent and re-installing it later is often very difficult!
Please refer to the layout diagram and PCB tracks diagrams below, and follow the steps carefully.
Use of a good quality soldering iron and solder is highly recommended for best results!
The following diagrams show the PCB layout and track diagrams of the QDX.
Tracks shown in BLUE are on the bottom layer. Tracks shown in RED are on the top layer. There
are only two layers (nothing is hidden in the middle). Not shown in this diagram are the extensive
ground-planes, on both sides of the board. Practically everything on both layers that isn’t a RED or
BLUE track, is ground-plane! The two ground-planes are connected at frequent intervals (not morethan 0.1-inches) by vias.
NOTE: the capacitor lead spacing on the PCB is 0.1-inches (2.54 mm) and most of the capacitors
are sized appropriately for this. From time to time, due to availability constraints, we may have touse capacitors with 0.2-inch lead spacing (5.08 mm); this is not a mistake, it is just due tocomponent availability. In this case simply use a pair of long-nosed pliers (etc) to straighten out
the wires and make them spaced for the 0.1-inch pads.
NOTE: PCB Rev 3a is different from Rev 3 only in that the trifilar transformer T2 was moved
slightly further from the PCB edge, and some components nearby were moved slightly too. Rev 3a
has a short-circuit from a via to ground, which must be resolved – see section 2.3a below.

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QDX assembly Rev 1.18d

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QDX assembly Rev 1.18d

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2.2 Parts list
Many components are SMD, pre-soldered to the PCB in the factory. Only through-holecomponents need to be installed by the constructor. SMD components in the parts list are
identified in the Description column and by the text colour being purple.
Resistors
Qty

Value

Description

2
2
6
2
7
4
7
4
4

0-ohms33-ohms
47-ohms
100-ohms
470-ohms
1K
10K
100K
470K

SMD 0603
SMD 0603
SMD 0603
SMD 0603
SMD 0603
SMD 0603
SMD 0603
SMD 0603
SMD 0603

Component numbers
R31, 32
R2, 4
R22, 23, 24, 25, 40, 41
R11, 34
R10, 12, 14, 26, 27, 28, 43
R1, 6, 20, 21
R3, 13, 15, 17, 18, 30, 35
R5, 19, 29, 42, 44
R7, 8, 9, 16

Capacitors (50V, Multi-layer Ceramic capacitors)(see below for listing of band-version dependent capacitors)
Qty

Value

Description

2
4
3
26

47pF
4.7nF
10nF
0.1uF

SMD 0603
SMD 0603
SMD 0603
SMD 0603

2
11
2
4
1
1

1uF
2.2uF
10uF
10uF
22uF
470uF

SMD 3216 tantalum
SMD 0603
SMD 0603
SMD 3216 tantalum
SMD 0405 electrolytic
SMD 0607 electrolytic

Component numbers
C27, 32
C44, 45, 59, 61
C36, 58, 60
C15, 16, 17, 21, 22, 23, 24, 25, 26, 33, 35, 37, 46, 47,
49, 50, 51, 52, 53, 54, 57, 66, 71, 72, 73, 74
C63, 69
C1, 2, 3, 4, 8, 34, 38, 42, 48, 50, 55
C64, 68
C62, 65, 67, 70
C36
C49

Semiconductors
Qty
4
1
6
1
1
2
1
1

Description
SMD: 1N4148 SOD323
SMD: SS14 DO-214AC
1N4007 diode
SMD: AMS1117-3.3
SMD: 78M05
SMD: FST3253 SOIC
SMD: 74ACT08
SMD: 24C64

Component numbers
D7, 8, 9, 10
DD1, 2, 3, 4, 5, 6
IC1
IC2
IC3, 4IC5IC6

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Qty
2
1
1
1
1
2
6
4
1
1
1
1

Description
SMD: LM4562
STM32F401RBT6
PCM1804
SMD: Si5351A / MS5351M
SMD: SN74LVC1G00DBVR
SMD: BSS84 MOSFET
SMD: BSS123 MOSFET
BS170: TO92 MOSFET
SMD: BC817 NNP SOT23-3
SMD: AOD403 (TO252)
3mm Red LED
SMD: 25MHz TCXO module

Component numbers
IC7, 9
IC8
IC10
IC11
IC13
Q1. 12Q2, 3, 4, 5, 6, 13
Q8, 9, 10, 11
Q14
Q7
PANEL
TCXO

Inductors(see below for listing of band-version dependent inductors)
Qty
5
1
4
1
1
1

Description
SMD: 47uH
SMD: 330uH (CD54)
47uH axial inductor
FT37-43 10t
BN43-202 binocular, 3:3
FT37-43 10t trifilar

Component numbers
L11, 13, 15, 16, 18
L17
L1, 5, 7, 9
L14
T1
T2

Miscellaneous
Qty

Value

1
1
1
1
1
1

2.1 mm DC
USB B
BNC
3.5mm jack
PCB
310 cm (80-20m version)
110 cm (20-10m version)
50 cm (80-20m version)
175cm (20-10m version)
M3 12 mm
M3
M3 12 mm1
1
1
1

Description
2.1 x 5.5 barrel DC power connector
USB type B connector
RF connector
3.5mm Stereo connector (PTT)
Main PCB, 86 x 59.5 mm
0.33 mm diameter wire (AWG #28)
0.60 mm diameter wire (AWG #22)
Steel 12mm long M3 screw
Steel M3 nut
Steel 12mm diameter M3 washer

QDX assembly Rev 1.18d

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Qty

Value

Description

2
1
1
8
4

Top, Bottom
Front panel
Rear panel
M2.5 machine screw
Rubber footExtruded aluminium top and bottom coverLaser etched, drilled for 3mm LED
Laser etched, drilled for three connectors
Screws to secure end panels
Self-adhesive rubber footFrequency dependent parts:
80 / 60 / 40 / 30 / 20m version of QDX:Capacitors (50V, Multi-layer Ceramic capacitors):
Qty

Value

1
1
2
1
1
1
1
1
2
2
2
1

22pF
30pF
56pF
82pF
100pF
180pF
220pF
270pF
390pF
470pF
820pF
1200pF

Description
Label “220”Label “300”Label “56J”
Label “820”Label “101”Label “181”Label “221”Label “271”Label “391”Label “471”Label “821”Label “122”Component numbers
C29
C31
C10, 28
C11
C9
C20
C30
C7
C6, 19C5, 14
C13, 18
C12

Inductors (powdered iron Micrometals toroids):
Qty
1
1
1
1
1
1
1

Description

Component numbers

T37-2 toroid (red), 2.40 uH (24t) L2
T37-2 toroid (red), 2.88 uH (26t) L6
T37-6 toroid (yel), 1.06 uH (18t) L3
T37-6 toroid (yel), 393 nH (11t) L4
T37-6 toroid (yel), 1.20 uH (20t) L8
T37-6 toroid (yel), 525 nH (13t) L10
T50-2 toroid – tapped, see text L12

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20 / 17 / 15 / 12 / 11 / 10m version of QDX:Capacitors (50V, Multi-layer Ceramic capacitors):
Qty

Value

Description

2
1
1
1
1
3
1
1
3
1
1

30pF
33pF
39pF
15pF
22pF
120pF
47pF
82pF
270pF
180pF
560pF

Label “300” or “30J”
Label “330” or “33J”
Label “390” or “39J”
Label “150” or “15J”
Label “220” or “22J”
Label “121”Label “470” or “47J”
Label “820” or “82J”
Label “271”Label “181”Label “561”Component numbers
C29, 31
C28
C30
C11
C9
C5, 18, 20
C10C7
C6, 12, 19
C14
C13

Inductors (powdered iron Micrometals toroids):
Qty
1
1
1
1
1
1
1

Description
T37-10 toroid (shiny black), 640 nH (16t)T37-10 toroid (shiny black), 722 nH (17t)T37-10 toroid (shiny black), 202 nH (9t)T37-10 toroid (shiny black), 250 nH (10t)T37-10 toroid (shiny black), 250 nH (10t)T37-10 toroid (shiny black), 302 nH (11t)
T50-6 toroid – tapped, see text

QDX assembly Rev 1.18d

Component numbers
L2
L6
L3
L4
L8
L10
L12

12

2.3 Inventory parts
Note: Photo shows parts for PCB Rev 3, 80/60/40/30/20m version of the QDX.
Note: Capacitor colours vary depending on availability. All are NP0/C0G Class-I RF dielectric.

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2.4 Install all the ceramic capacitors
80 / 60 / 40 / 30 / 20m version: (see following page for high-band QDX version): Install all 16
through-hole capacitors in accordance with the following diagram. The procedure is so easy that I
have included all capacitors in a single assembly step rather than doing one diagram for eachcapacitor value. Nevertheless be very careful to insert the correct value capacitors in the correct
places. Mistakes are hard to correct later.
In the diagram, the component label (capacitor body inscription) is written inside the capacitor
body, which may be supplied coloured yellow, or blue, according to availability. The actual
value e.g. 220pF is written in red text next to the capacitor. Note that the leads of some capacitors
may need to be bent to fit the 2.5mm holes.
Qty

Value

2
2
1
1
2
1
1
2
1
1
1
1

470pF
390pF
270pF
100pF
56pF
82pF
1200pF
820pF
180pF
22pF
220pF
30pF

Description
Through-hole, label “471”Through-hole, label “391”Through-hole, label “271”Through-hole, label “101”Through-hole, label “56J” or “560”Through-hole, label “820”Through-hole, label “122”Through-hole, label “821”Through-hole, label “181”Through-hole, label “220”Through-hole, label “221”Through-hole, label “300”Component numbers
C5, 14
C6, 19C7C9C10, 28
C11
C12
C13, 18
C20
C29
C30
C31

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20 / 17 / 15 / 12 / 11 / 10m version: Install all 16 through-hole capacitors in accordance with thefollowing diagram. The procedure is so easy that I have included all capacitors in a single
assembly step rather than doing one diagram for each capacitor value. Nevertheless be very
careful to insert the correct value capacitors in the correct places. Mistakes are hard to correct
later.
In the diagram, the component label (capacitor body inscription) is written inside the capacitor
body, which may be supplied coloured yellow, or blue, according to availability. The actual
value e.g. 220pF is written in red text next to the capacitor. Note that the leads of some capacitors
may need to be bent to fit the 2.5mm holes.
Qty

Value

Description

2
1
1
1
1
3
1
1
3
1
1

30pF
33pF
39pF
15pF
22pF
120pF
47pF
82pF
270pF
180pF
560pF

Label “300” or “30J”
Label “330” or “33J”
Label “390” or “39J”
Label “150” or “15J”
Label “220” or “22J”
Label “121”Label “470” or “47J”
Label “820” or “82J”
Label “271”Label “181”Label “561”Component numbers
C29, 31
C28
C30
C11
C9
C5, 18, 20
C10C7
C6, 12, 19
C14
C13

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2.5 Install 1N4007 diodes
Install the six 1N4007 diodes D1-D6.
Pay attention to the orientation of the diodes which is critically important. The diodes have a black
body and a white stripe. The white stripe end must be aligned with the white stripe on the image
on the PCB silkscreen. In the diagram below, all the white stripe ends of the diodes are on the leftof each diode position.

So far, the assembly
should look like this →(80 / 60 / 40 / 30 / 20m Rev
4 version pictured)

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2.6 Install 47uH inductors
Install the four 47uH axial inductors L1, L5, L7 and L9. Bend one lead over 180-degrees so thatthe inductor body stands up vertically at 90-degrees to the PCB.
The orientation of these is not particularly important, but my recommendation is to follow thesilkscreen PCB pattern, putting the body (longer lead) nearer to the PCB lower edge (as orientatedin the diagram below) and the wire (shorter lead) nearer to the top side.
NOTE: Several people worried that there
should not be a connection at the end of L1
nearest the board edge, to the thick output
trace. There IS a connection here! So don’t
try to remove the connection, a solder
bridge between one end of L1 and one endof D3 is normal and CORRECT!
(photo by Paolo 31JS20)

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2.7 Install PA transistors
Install the transistors in the positions shown, with their flat faces flush against the PCB.
Use the 12mm M3 bolt, washer and nut to firmly press the transistor faces against the PCB, as
shown.
Refer to the diagram and photographs below.

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2.8 Assemble and install transformer T1
Transformer 1 is wound on the BN43-202 binocular ferrite former, using the thick 0.6mm (AWG
#22) wire. This wire is also used for L14 so do not use all of it on the transformer.
It is a good idea to de-burr the toroid because sharp edges can damage the insulating enamel on
the wire – however be careful since the ferrite is brittle and can easily be broken. So GENTLE
scraping of the edges with something sharp like a knife is possible.
The transformer has two windings, both 3 turns (3:2 if building for 12V supply). The three-turnprimary winding has a center-tap. In the nomenclature of binocular cored transformers, “1 turn”
means the wire passes through both sets of holes, ending up back at the end where it started.

Assembly of this transformer is best done in steps as follows. Please read all the steps before
commencing the assembly:
Step 1: Carefully unwrap the thick 0.6mm (AWG #22) wire, and straighten it ensuring there are nokinks.
Step 2: Pass the wire through both holes, starting at the top left, as shown. This is the first turn of
the 3-turn primary winding.
Step 3: Now pass the wire through the top hole of the binocular core, back from left to right. This is
the next half-turn of the primary winding, bringing the number of turns so far to 1.5; now we must
make the center tap.

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Step 4: Now pass the wire back through the bottom hole of the binocular core, from right to left;
but do not pull it tight. Leave a small loop as shown in the photograph. This will be soldered into
the center-tap pad on the PCB.
Step 5: Wind the wire through both the top hole of the binocular core and the bottom hole, pulling
it a little tight as normal; this forms the final turn of the 3-turn primary winding. You end up with thewire coming out of the bottom left hole as shown; cut it to about 1cm protrusion.

Step 6: Squeeze together the center-tap to try to avoid subsequent confusion.
Step 7: Start the 3-turn secondary winding by pushing the wire from right to left through the bottom
hole of the binocular core, then from left to right through the top hole. This is the first turn.

Step 8: Push the wires through both holes twomore times to create the second turn: right to left
through the bottom hole, then left to right throughthe top hole. Now we have three turns on thesecondary. Cut the wire leaving about 1cm spare.
Remember to wind only 2 turns secondary, if
winding the transformer as 3:2 for 12V
operation.
Please refer to page 4 for details of 9V or 12V
operation choices.

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Step 9: Bend all the wires to point downward in
the position they will need to be, to fit through theholes in the PCB.
Step 10: Now cut the center-tap loop; hopefully,
as mine are in this photograph, these two wires
formed by cutting the loop are LONGER than thetwo ends of the 3-turn secondary, so that you
cannot mix anything up.
Step 11: Insert the wires into the holes on the
PCB. The two wires (of the cut loop) that belong
to the center-tap of the 3-turn primary, which are
the longer wires, are inserted into the middle holeon the right-hand side; this hole is larger than the others and is big enough for both wires to fit in.

Step 12: Now we need to get the enamel off
the wire. Usually on thinner wire, I hold the soldering iron to the wire until the enamel burns off. Butthat doesn’t work so well on thicker wire such as this. So my technique here is to scrape theenamel off, at least partially, using wire cutters. The correct pressure needs to be applied to thewire cutters, so as NOT to actually cut the wire. I hold the wire cutter as close to the PCB as
possible, then gently but firmly pull the cutter away from the PCB, scratching off the enamel. Turnthe cutter to a different angle and scrape again, 2 or 3 times. It is not necessary to remove ALL theenamel, if you get a few good scrapes on, enough enamel will be removed that the soldering ironheat will burn off the rest of the enamel and a good joint will be achieved. Do this for the 4 winding
ends first, leave the two center-tap for later, to make it easier.
Step 13: Cut each of the four wire-endings, leaving about 2mm or less, sticking up from the PCB.

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Step 14: Solder the four wire endings. At each pad, hold the soldering iron firmly, apply plenty ofsolder, and wait for about 10 seconds. This will ensure a good connection, and any remaining
enamel will be burned away.
Step 15: Finally repeat the procedure with the two center-tap wires that came through the large
center hole. Scrape them, cut them to 2mm, and solder them. Apply plenty of solder and hold thesoldering iron tip to the joint for maybe 15 seconds, to really be sure of a good joint and burningaway any remaining enamel.

Step 16: Check that none of the wires protrude more than 2mm from the surface of the PCB, since
if you are using the optional aluminium enclosure, there are only a few mm clearance.
Step 17: Verify good connections for all five soldered joints of T1 using a DMM set to resistance orcontinuity mode. My cheap DMM hasn’t got a continuity mode and I’m using the 2000-ohm
resistance mode; in this mode when there is continuity, the reading on mine shows 001 (not zeroohms; this is just a DMM thing, of no significance).

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You should see continuity between all the points labeled A in the photo below. Touch the probes
to pairs of these points, to verify continuity.
You should see continuity between the points labeled B also. Touch the two probes to the twopads labeled B and check for continuity.
Finally, there should be NO connection between A and B. Hold one probe on any A point and theother probe on a B point. That should read infinite resistance (no continuity).
If any of these tests fail, then you have a soldering problem somewhere, or the wrong wire in thewrong hole, or some short-circuit somewhere etc.

This is the final appearance of T1 when the installation is complete:

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2.9 Prepare and install tapped inductor L12
80 / 60 / 40 / 30 / 20m version (high bands version see next section): L12 is an inductor wound
on a T50-2 toroid (red colour, large size), having several taps which are switched in according tothe operating frequency (band). There are 41 turns in total, with taps at 19, 30 and 36 turns. The
diagram shows the location of the taps at 19, 30 and 36 turns, which have larger holes, such that
two wires can be inserted.
Carefully unwind and cut off 75cm of the 0.33mm (AWG #28) enameled wire, and straighten it,
ensuring no kinks.

Step 1: Begin winding L12 as shown. Remember that in the nomenclature of toroidal inductors,
each time the wire passes through the center of the toroid counts as 1 turn.
Step 2: Wind 19 turns then make a loop between the 19 th and 20th turns. This is for the 19 turn tap.

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Step 3: Wind up to the 30th turn then make another loop between the 30th and 31st turns. This is forthe 30 turn tap.
Step 4: Wind up to the 36th turn then make another loop between the 36th and 37th turns. This is forthe 36-turn tap.

Step 5: Wind the remaining turns to complete the total of 41 turns.

Step 6: Now with a pair of pliers, you can squeeze each of the loops into a sharp
point, such that it is easy to fit through the large holes at the 19, 30 and 36-turntaps (see photo, right).
Step 7: Insert all the wires in theircorrect holes.
You should pull them firmly from theunderside of the board to ensure that
there are no loose wires on the top side.
Now cut each wire to about 2mm length(at most) and solder in place. It isessential to remove the enamel fromthe wire. My favourite method of doing
this is simply to hold the soldering ironto the joint, with plenty of solder, for atleast 10 seconds. The enamel burns
away in this time.

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Step 8: Verify the joints are properly soldered, by using a DMM in continuity testing mode (if it has
this mode) or check for zero ohms in resistance mode. On the reverse of the PCB, check forcontinuity between pins 3, 4, 5, 6 and 9 of IC3 as shown. You should measure 0 ohms (continuity)
between any pairs of these. If you do not, then there is a mistake somewhere, most probably a
failure to burn away the enamel at one or more of the L12 connections to make a good joint.

Here’s the story so far, including the nicely installed L12 gtapped inductor.

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20 / 17 / 15 / 12 / 11 / 10m version: L12 is an inductor wound on a T50-6 toroid (yellow colour,
large size), having several taps which are switched in according to the operating frequency (band).
There are 20 turns in total, with taps at 11, 14 and 17 turns. The diagram shows the location of thetaps at 11, 14 and 17 turns, which have larger holes, such that two wires can be inserted.
Carefully unwind and cut off 50cm of the 0.33mm (AWG #28) enameled wire, and straighten it,
ensuring no kinks.

Step 1: Begin winding L12 as shown. Remember that in the nomenclature of toroidal inductors,
each time the wire passes through the center of the toroid counts as 1 turn.
Step 2: Wind 11 turns then make a loop between the 11 th and 12th turns. This is for the 11 turn tap.

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Step 3: Wind up to the 14th turn then make another loop between the 14th and 15th turns. This is forthe 14 turn tap.
Step 4: Wind up to the 17th turn then make another loop between the 17th and 18th turns. This is forthe 17-turn tap.

Step 5: Wind the remaining turns to complete the total of 20 turns.

Step 6: Now with a pair of pliers, you can squeeze each of the loops into a sharp
point, such that it is easy to fit through the large holes at the 11, 14 and 17-turntaps (see photo, right).
Step 7: Insert all the wires in theircorrect holes.
You should pull them firmly from theunderside of the board to ensure that
there are no loose wires on the top side.
Now cut each wire to about 2mm length(at most) and solder in place. It isessential to remove the enamel from thewire. My favourite method of doing thisis simply to hold the soldering iron to thejoint, with plenty of solder, for at least 10
seconds. The enamel burns away in thistime.

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Step 8: Verify the joints are properly soldered, by using a DMM in continuity testing mode (if it has
this mode) or check for zero ohms in resistance mode. On the reverse of the PCB, check forcontinuity between pins 3, 4, 5, 6 and 9 of IC3 as shown. You should measure 0 ohms (continuity)
between any pairs of these. If you do not, then there is a mistake somewhere, most probably a
failure to burn away the enamel at one or more of the L12 connections to make a good joint.

Here’s the story so far, including the nicely installed L12 gtapped inductor.

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2.10 Wind and install L14
L14 consists of 10 turns of 0.6mm (AWG
#22) wire wound on an FT37-43 toroid(dull black colour). Firstly, you have tounderstand that there’s a right way and awrong way to wind a toroidal inductor,
too. The terms “right” and “wrong” areprobably not as appropriate here, as“handedness” or “winding direction”.
There are two directions you can wind
the toroid. If you choose the right one,then all the wires will automatically end
up near the holes they are to be soldered
into. If you do it wrongly, it will be a bit
messy.
For all the toroidal inductors in the QDX kit, you will get it right, if you start as shown (photo, above
right) and pass the wire through the toroid from the top side down through the hole, out and thenaround and over again; accumulating turns in the anticlockwise direction.
With this in mind, wind 10 turns and install the toroid. As with the output transformer T1, I
recommend scratching the enamel with a knife or wire-cutter, and then cutting it to about 2mm
length on the underside of the board. Then solder, applying the soldering iron for at least 10
seconds and plenty of solder, to ensure a good connection. Do NOT tin the wires before inserting
into the holes: the holes are not large enough to accommodate tinned wires.

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2.11 Install Low Pass Filter toroids
80 / 60 / 40 / 30 / 20m version: Now we’ll wind and install thesix Low Pass Filter toroids using 0.33mm (AWG #28) enameledwire. Remember, as per the previous direction, using the right
winding direction will make the toroids much easier to fit to the
PCB. Refer to the table below. Make sure to count carefully andinstall the toroids in the correct places! Remember that eachtime the wire passes through the center of the toroid counts as
one turn. Spread the turns evenly with a small gap.
Winding the toroids as tightly as possible will slightly improve
the performance of the Low Pass Filters. But don’t pull sotightly that you break the wire!
The following table lists the required number of turns, cut wire
length, and inductance.
Band

Part number

80m

L2

80m

Core

Inductance

Number of turns

Wire lengthT37-2 (RED)

2.40uH

24

36cm

L6

T37-2 (RED)

2.88uH

26

39cm

60 / 40m

L3

T37-6 (YELLOW)

1.06uH

18

29cm

60 / 40m

L8

T37-6 (YELLOW)

1.20uH

2030cm

30m/20m

L4

T37-6 (YELLOW)

393nH

11

20cm

30m/20m

L10

T37-6 (YELLOW)

525nH

13

22cm

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20 / 17 / 15 / 12 / 11 / 10m version: Now we’ll wind andinstall the six Low Pass Filter toroids using 0.60mm (AWG
#22) enameled wire on the shiny black toroids. You willneed to scrape the enamel off the wire and tin the copper
before installation. Remember, as per the previous direction,using the right winding direction will make the toroids mucheasier to fit to the PCB. Refer to the table below. Make sure
to count carefully and install the toroids in the correct places!
Remember that each time the wire passes through the centerof the toroid counts as one turn. Spread the turns evenly witha small gap.
Winding the toroids as tightly as possible will slightly improve
the performance of the Low Pass Filters.
The following table lists the required number of turns, cutwire length, and inductance.
Band

Part numberCore

Inductance

Number of turns

Wire length

20m

L2

T37-10 (BLACK)

640nH

16

25cm

20m

L6

T37-10 (BLACK)

722nH

17

27cm

17 / 15m

L3

T37-10 (BLACK)

202nH

9

16cm

17 / 15m

L8

T37-10 (BLACK)

250nH

10

18cm

12 / 11 / 10m

L4

T37-10 (BLACK)

250nH

10

18cm

12 / 11 / 10m

L10

T37-10 (BLACK)

302nH

11

20 cm

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2.12 Wind and install trifilar toroid T2
This toroid needs some care so please follow these instructions very carefully.
Divide the remaining 0.33mm (AWG #28) wire into three equal lengths. These three pieces now
need to be tightly twisted together to make the trifilar wire. My method for this is to tie one end in aknot around a small screwdriver shaft. Similarly tie the other end around another small
screwdriver. Now clamp one end somehow to something solid. You could use a vise, if you haveone. If you don’t, then you have to get creative and think of something. Here I taped it to the edgeof the desk. Now you can twist the screwdriver at the free end, repeatedly until you twist the threewires together thoroughly. You need to keep the wire under a little tension to keep the twists
evenly spaced.

A 20cm length of wire should have about 60 twists. It is not critical.
The end result is something like the photo (right). The measurement scale is in cm.
Now cut off the untidy ends, and this is the piece of wire that will be used to wind the FT37-43
toroidal core as a trifilar transformer.

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Hold the core between thumb and finger. Pass the wire first from above, to below. Then take thewire from below, and bring it around to pass through the toroid again to form the second turn. Aftereach turn, ensure the wire is fitting snugly around the toroidal core. Wind 10 turns on the core.
Each time through the toroid’s central hole counts as one turn. Cut off the excess wire, leaving
about 2.5cm remaining.

Now it’s necessary to identify which wire belongs to which winding. You have three windings
twisted together, they all use the same wire. The only way to do this is with a DMM as continuity
tester. First, untwist and straighten the wire ends that are not wound around the toroidal core.
Now tin the last few mm at the ends of each wire. You can do this by scraping off the enamel thentinning with the soldering iron; or, if your soldering iron is powerful enough, hold the wire end in ablob of molten solder for a few (maybe 10) seconds, until the enamel burns off.
Now use a DMM to test forcontinuity. Re-arrange the wires
so that there is continuity from
A-A, B-B, and C-C in this photo.
Carefully keep this orientation ofwires and insert the transformer
this way into the PCB. BE
CAREFUL not to lose theorientation of the wires! The
right wires must be in the rightholes, so that the windings areconnected correctly in thecircuit!
Refer to the diagram on the following page to see the correct orientation.
Once the wires are inserted through the correct holes in the PCB, and pulled tight, check forcontinuity between the pairs of wires in holes A, B and C once AGAIN. It is much easier to get thisright first time, than it is to make repairs later!
Be sure NOT to solder any wire into the hole marked “GND”...
When you are satisfied that the wires are all in the correct holes, you can cut them to a length ofabout 2mm and solder them. It is best to cut-and-solder one wire at a time, since if you cut all the

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wires to 2mm length then the toroid is more likely to fall out before you’ve had a chance to solder
any of the wires. If that happens, it will be tough to get all the wires back in the correct holes again.

There are numerous SMD components in the vicinity so be VERY careful when soldering the wiresof toroid T2, NOT to touch any of the nearby SMD components!
Once again my method
involves holding the ironand plenty of solder on the
joint for at least 10
seconds to make sure theenamel burns offcompletely.
The photograph (right)
shows the trifilar wire
ending holes labeled A B
C on the lower side of the
PCB. You should checkfor continuity between
each pair of pins A-A, B-B
and C-C; and the triplet ofpins IC4 pins 7 and 9, andIC3 pin 7.

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2.13 Install the 3mm red status LED
Identify the shorter of the two wires of the 3mm red LED and bend it 90degrees right at the body of the LED.
Bend the other wire 90-degrees but at a distance of approximately 2.5mm(0.1-inches) from the body of the LED (see photograph, right).
Insert the LED leads into the holes in the PCB. As an additional check, theLED will have a flat part on its body, which will sit flat on the surface of the
PCB. Solder the LED In place but be careful that it is accurately aligned inposition, protruding over the board edge at right-angles. This will ensure thatlater if you are using the optional enclosure, the LED will fit without difficulty.

Photo (right) shows the LED installed
correctly (Rev 3 PCB pictured)

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2.14 Install jumper wire
For the 20 / 17 / 15 / 12 / 10m version only:
The jumper wire installation is necessary to inform the firmware to set up the band configuration
parameters according to the correct defaults for the high-bands 20-10m version.
Just to the left of the BS170 power amplifier transistors is the QRP Labs Logo. Use a piece ofcomponent wire off-cut formed into an inverted ‘U’, and install it in the two holes which are abovethe ‘b’ and ‘s’ of “Labs”, as shown in the photograph below. Be sure to ignore the hole which is at
the bottom of the ‘P’ of “QRP”.
With this jumper wire installed, QDX will automatically initialize itself for high-bands operation.

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2.15 Install connectors
The final components to install are the connectors, from top right to bottom right:••••

X1: USB B connector
X2: DC Power connector
X2: PTT (mis-named X2, should be X3)
X4: BNC RF connectors – NOTE: there is a footprint for an SMA connector that you may
install if you wish; however this will have gaps around it as it pokes through the enclosure.

It is very important to install these carefully aligned so that they are straight and at right-angles
with respect to the PCB. The connector body should not protrude beyond the edge of the PCB.
When the connectors are well-aligned, you will have no trouble fitting the PCB into the optional
enclosure. Clip off excess leads and tabs to make sure nothing protrudes more than 1.5mm.

Assembly of the
QDX kit PCB isnow complete!
(Rev 4 PCB,
80/60/40/30/20m
version shown).

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2.16 Optional enclosure
Installation of the QDX PCB in the optional enclosure is very straightforward and self-explanatory.
1. Install the screws at the bottom left and bottom right of the front panel, fixing it to the bottom
half of the enclosure. Note that the enclosure top and bottom are identical.
2. If the short edges (front and back, i.e. LED status edge, and connectors edge) of the PCB
feel rough, it can be a good idea to gently file them so that the PCB edge is square.
3. Gently slide the PCB into the guide-rail grooves in the enclosure bottom.

4. Install the two screws at the bottom left and bottom right of the rear panel, fixing it to thebottom half of the enclosure.
5. Fit the top half of the enclosure. Note that one side has an “I-tongue” and the other side has
a “U-groove” and this is repeated on both halves (top and bottom) of the enclosure.
Therefore the top half will only fit the bottom half, “one-way-round”. If it does not appear tosit correctly, rotate the top half 180-degrees. Secure the top with the remaining screws.
6. Finally you may, if you wish, fit the nut to the BNC connector. I generally leave off thewasher. The nut may be omitted too, it is not really required and should in any event not be
over-tightened since there is a small gap between the back surface of the rear panel, andthe BNC connector body. The washer can be installed on BNC connector the INSIDE of theenclosure to fill some of that gap. It’s a matter of trying it out and seeing what fits best in
your case and gives you a result you’re most happy with.
7. If desired, attach the self-adhesive rubber feet to the bottom of the enclosure in the corners.

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QDX assembly Rev 1.18d

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3Design3.1 Summary
Traditionally it was normally assumed that transmission of digital modes requires a computer, togenerate the audio tones, and an SSB transceiver, to modulate those to RF and transmit them via
an antenna.
Yet – the widespread assumption that an SSB transceiver is REQUIRED is in fact absolutely
incorrect, at least for a wide category of digital modes transmitting a single tone frequency shift
keyed signal. This class of transmissions are NOT SSB, they are a single carrier, frequency
modulated. Potentially this can provide simplification of the radio transceiver design, deliveringboth higher performance and lower cost.
Why is this important? An SSB transmitter design is necessarily relatively complex and a nontrivial undertaking for several reasons. The audio signal must be converted to RF by an SSB
exciter. An exciter may commonly take the form of a superheterodyne architecture having a
modulator, IF filter (crystals) and another mixer to convert to the final RF operating frequency. Itthen requires a linear amplification chain to raise the signal amplitude to the required power output
at the antenna port. An alternative architecture is a phasing network exciter which makes use ofphasing techniques to null the unwanted sideband, and converts directly from baseband to RF.
Again it is followed by a linear amplification chain. In both cases considerable attention to thedesign is required, to minimize the residual carrier frequency, and the unwanted sideband.
Linearity must be maintained throughout the amplifier chain to the antenna port, to avoid
splattering onto adjacent frequencies. Even though the audio signal is theoretically a single audio
tone, harmonics of the audio tone may be present particularly if the audio level is too high, and thecarrier suppression and unwanted sideband are never perfect; so there is plenty of potential for
generation of intermodulation products, hence the need for good linearity throughout. A linear
amplifier (for example, Class AB) has low efficiency compared to non-linear amplifiers such as
Class C. Low efficiency means higher current draw for a given power output (less battery-friendly
in portable scenarios), and higher heat production, probably requiring heatsinking of the PA
transistors – increasing size, cost and weight.
Double Sideband (DSB) is even worse – the designs are simpler but here we have an equal
amplitude unwanted sideband emission. Half the transmitter’s power is wasted on this unwanted
sideband, which potentially can interfere with users of adjacent channels. On receive, we suffer a
3dB degradation in noise floor at best, and at worst, interference from signals in the unwanted
sideband. With the strong double tone (both sidebands) the potential for intermodulation products
due to non-linearities in the amplification chain is higher. Except under optimal conditions, andeven then with difficulty, two DSB transceivers cannot QSO with each other! A DSB transceiver
can only QSO with an SSB transceiver at the other end. DSB is the only mode which cannot evencommunicate with itself! Some would say, an abomination of a mode.
If there were a way to convert the audio tones from the computer, to RF, without going through the
SSB modulation process and linear amplification, it could solve many problems.
Back in 2012 the QRP Labs Ultimate QRSS/WSPR/Digi transmitter kit was the first kit to generate
and transmit digital signals standalone with no PC required, and without an SSB transmitter. It
used just a microcontroller calculating the required frequency shifts and commanding an RF signal
generator – then the AD9850 Direct Digital Synthesis (DDS) chip – to produce the required
frequencies. Several iterations of the kit evolved over the years, to the current Ultimate3S
http://qrp-labs.com/ultimate3/u3s which can transmit quite a list of digital modes in the standalone
beacon application.
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However for QSO digital modes, using computer (PC) software to encode and decode the audio,
SSB transceivers remained the norm.
In the second half of 2019, I was thinking about FT8, and a number of simple low cost FT8transceiver kits that were available (DSB transceivers), none of which offered a particularly good
performance. I realized that there IS a better way to do this, that will dramatically increase theperformance without increasing the price. Straight away I designed QDX and built the prototype.
The fact that the computer is generating the audio, does not need to stop us from generating asingle RF tone from an RF signal generator, just as the Ultimate3S QRSS/WSPR/Digi kit does. A
microcontroller can analyze the audio arriving from the PC, and determine the audio frequency of
the tone; then add that to an RF base frequency (which on an SSB radio, we would call the “USB
Dial Frequency”), to determine the correct RF carrier frequency to generate for transmission. The
rest is then a matter of commanding an RF generator to produce the calculated frequency, andamplifying it for presentation to the antenna. The generated signal is absolutely clean, having only
harmonically related unwanted spurious emissions which can be taken care of in the conventional
way by Low Pass Filtering.
The importance of this technique cannot be understated.
•

We no longer need a complex SSB modulator involving multiple mixers, IF filters (orphasing in the case of the direct method).

•

No more linear amplification chain – we can use higher efficiency non-linear amplifierclasses, reducing the need for heatsinking, and lowering transmit current, very battery
friendly.

•The unwanted sideband simply DOES NOT EXIST. It isn’t just “there but attenuated”
depending on how well the transceiver was designed and aligned - it just isn’t there at all.

•Similarly the carrier also does not exist. Carrier suppression in a modulator depends on
how well the mixer is balanced and is never perfect. But here, we generate a single tone
directly – and there’s NO SSB carrier at all. Zero.

•

It is impossible to overdrive this design by turning up the audio volume too high on the PC.
Therefore it is impossible to generate a messy signal that splatters on to adjacentfrequencies.

With this in mind, the rest of the QDX transceiver design falls into place, around this central
concept.

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QDX assembly Rev 1.18d

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3.2 Block Diagram

QDX consists of the following circuit blocks:
1. A synthesized Local Oscillator using the famous Si5351A Digital PLL signal generator chip.
A 0.25ppm 25MHz TCXO is included as standard as the PLL reference.
2. Low Pass Filtering to attenuate emissions at harmonics of the operating frequency. Filters
are provided to cover 80, 40, 30 and 20m bands, and selected using solid state, PIN diodeswitching.
3. A solid state transmit/receive switch – no bulky, noisy, expensive, unreliable relays here!
4. A switched receiver band pass filter to provide some degree of protection to the mixer
against out of band signals.
5. A high performance, double balanced Quadrature Sampling Detector to mix the incoming
RF signals to baseband and produce I and Q signals.
6. A low noise pre-amplifier for the I and Q signals.
7. High performance 24-bit 48ksps (Kilo Samples Per Second) stereo Analog To Digital
Converter (ADC) chip with 112dB of dynamic range.
8. An embedded Software Defined Radio (SDR) receiver which implements, digitally, a 12kHzIntermediate Frequency superheterodyne receiver with excellent performance andunwanted sideband suppression, also implementing a sharp digital filter.
9. An embedded 24-bit 48ksps stereo USB Sound card – no more noisy, cluttered audio
cables!
10. A CAT control serial interface, also over the same USB cable, to allow the PC software tocontrol the radio (frequency, transmit/receive switching etc) in a standard way.
11. The microcontroller performs single cycle frequency analysis of the audio tone, andcommands the Si5351A Signal generator to produce the necessary RF frequency.

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12. A Class-D Push-Pull power amplifier that is small, low cost, high efficiency, and producesvery low even harmonic output levels, reducing the demands on the Low Pass Filter.
13. Voltage regulation and supply decoupling.
In subsequent sections, each of these blocks will be described in detail.

3.3 Synthesized local oscillator
Generating a stable, precise oscillator signal was one of the most challenging aspects oftransceiver design. Modern semiconductors however, make it one of the easiest building blocks of
the radio.
The synthesizer used in the QDX is the same as in the QCX 5W CW transceiver, and various
other QRP Labs products: the Si5351A. This is a Digital Phase Locked Loop (PLL or DPLL)
synthesizer which provides three separate frequency outputs, each having a frequency range
spanning 3.5kHz to 200MHz. The frequency stability is governed by the a crystal reference.
Due to unavailability of the Si5351A, theequivalent MS5351M may be used. For test
details on the performance of Si5351A vs
MS5351M demonstrating the suitability of theMS5351M (in fact, slight superiority in many
regards), please see
http://qrp-labs.com/synth/ms5351m.html
The block diagram (right) is taken from the
SiLabs Si5351A datasheet. Briefly, the 27MHz
reference oscillator is multiplied up to an internal
Voltage Controlled Oscillator in the range 600-900MHz (the PLL), then divided down to produce
the final output frequency. The multiplication up and the division down are both fractional and so
the frequency resolution is extremely finely controlled. The chip has two PLLs and three output
divider units.
For best jitter performance, the Si5351A datasheet recommends the use of even integer dividers(no fractional component) in the MultiSynth dividers and in the QDX transceiver design, thisrecommendation is followed.
The Si5351A datasheet dictates the use of a 25 or 27MHz crystal or oscillator. In the QDX a
25MHz crystal was originally used, together with an unusual NPN/PNP transistor oscillator circuit,which produces a near rail-to-rail (3.3V peak-peak) output that is perfect for driving both theSi5351A and the STM32 processor. However on Rev 3 of the PCB, layout changes caused highinductive loading on the 25MHz signal. During transmit the synthesizer was found to suffer
momentary loss of PLL lock, causing discontinuities in the transmitted signal which sounded like“pops” when monitored on a different radio receiver.
The “popping” was fixed via a firmware update (1_04) however this effectively hid the pops, rather
than cured the root cause. In 1_04 the “pops” are so infrequent as to be virtually undetectable.
The actual root cause is that the Si5351A is very sensitive to the waveform driving it on pin 2.
According to the datasheet, a 1Vpp signal should be used. However in practice this is found to beincorrect; a larger signal is frequently required, to ensure no loss of lock (“pops”). In Rev 4 of the
PCB, a single NAND logic gate chip is used to buffer the 1.1Vpp signal from the 25MHz TCXO toa 3.3V squarewave. This completely cures the “pops”.
QDX assembly Rev 1.18d

45During testing, the original 25MHz push/pull crystal oscillator was found to provide an
unsatisfactory level of frequency stability, which was very disappointing and boards had alreadybeen ordered. In the graph below, an example operating session is shown. QDX was operated on
the 20m band and the operating frequency was monitored as an audio tone on a separate
receiver. 20m was used as it is the highest frequency of operation of QDX and would therefore
illustrate the worst-case scenario. The nominal operating frequency was 1kHz audio and was notaltered during the test.
In the graph, the red line is the audio frequency. The blue line is the output of the STM32’s internal
temperature sensor ADC input. It is uncalibrated (to temperature), and just indicates relative
temperature changes.

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The QDX was switched on at 00:00 on the left side of the graph. An initial warm-up drift of 30Hz
downward drift can be seen (red line). At around 12 minutes, FT8 operations were commenced,
having the usual 15 second slots with alternating transmit and receive cycles, resulting in a duty
cycle a little under 50% (FT8 transmissions are a bit less than the 15 second cycle). FT8
operations ceased at around 47 minutes, by which time the enclosure had warmed up
considerably and the frequency had dropped almost 90 Hz from power-on! As the unit cooled
down, the frequency drifted back upwards again. The temperature curve (blue) gives a relative
indication of the temperature of the PCB.
FT8 has a bandwidth of about 50Hz. You can see that a 90Hz drift in half an hour of operation israther unacceptable! Anybody who has tried FT8 operation will know that the band is usually full of
QSOs and you struggle to find a clear spot to transmit in, often slotted in with other stations closelyon either side. With such a bad drift rate, you would drift right across the other stations’transmissions – or to avoid that, have to keep making little adjustments to the operating audio
frequency in WSJT-X to counteract the drift! Horrible!
The problem was resolved in the first QDX batch by designing a neat little PCB containing a
25MHz TCXO that is fit to the QDX during assembly, INSTEAD of the 25MHz crystal.
In subsequent PCB revisions the 25MHz TCXO is already mounted on the main PCB.
The output amplitude of the TCXO is 1.1V peak-peak which is enough for the Si5351A, but
insufficient to drive the STM32 oscillator input. It was therefore buffered using the original
NPN/PNP transistor crystal oscillator circuit, which resulted in a rail-to-rail (3.3V peak-peak) nearsquare wave.
After fitting the TCXO in place of the original 25 MHz crystal, the drift is reduced to +/- 1Hz or sowhich is unnoticeable and entirely acceptable.
In summary:
PCB Rev 1:

NPN/PNP 25MHz crystal oscillator suffered frequency drift, and a retrofit 25MHzTCXO daughtercard was supplied, fitted in place of the 25MHz crystal, to solve thedrift problem. The same NPN/PNP push-pull circuit now functioned as a signal
buffer.

PCB Rev 2:

The 25MHz TCXO is fitted as standard on the PCB (no longer a daughtercard).

PCB Rev 3:

PCB layout changes to accommodate the PCM1804 ADC chip and differential
drivers was thought to result in increased parasitic capacitance and inductance,loading the 25MHz signal and reducing the signal amplitude slightly. Random lossof PLL lock occurred, causing “pops” when the signal was monitored on a receiver;firmware 1_04 hid this problem by finding different ways to configure the Si5351A
that made PLL lock loss very rare.

PCB Rev 3a: As per Rev 3 above.
PCB Rev 4:

NPN/PNP buffer circuit changed to a singe logic gate NAND chip buffer, resulting inclean 3.3Vpp squarewave that completely solves the “popping” issue at source.

The Si5351A has a large number of internal 8-bit registers to control the synthesizer configurationand output frequency, and these are programmed by the microcontroller using the I2C serial
protocol. 1K resistors R20 and R21 are pull-ups required for the operation of the bus at 400kHz.
There are three outputs of the Si5351A synthesizer and these are all used to good advantage. The
Clk0/1 outputs are used to drive the Quadrature Sampling Detector (QSD) during receive.

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A feature of the Quadrature Sampling Detector is that either the RF input, or the LO input, must
provide two paths in 90-degree quadrature. This is normally applied at the Local Oscillator where it
can be easily controlled for best performance. So, two oscillator signals are required, with thesame frequencies but a precise 90-degree phase offset. Generating this quadrature Local
Oscillator signal is always difficult. Analogue phase shift circuits have limited accuracy. Often adivide-by-4 circuit is used, to produce quadrature oscillator outputs from an oscillator input at 4x
the reception frequency. This also creates challenges particularly as you try to increase thereception frequency to cover higher bands. For example, on 10m e.g. 30MHz, a local oscillator at
120MHz is required and the divide-by-4 circuit must be able to operate at such a high frequency.
Devices such as the 74AC74 can do so, but pushing it higher into the 6m band cannot be donewith the 74AC74.
The Si5351A has a phase offset feature, which is not really very clearly described in the SiLabs
documentation. However, QRP Labs has perfected the technique to put two of the Si5351A
outputs into precise 90-degree quadrature, which is maintained without tuning glitches as thefrequency is altered. It’s a nice development because it eliminates one more circuit block (the74AC74 divide-by-4 circuit), again reducing complexity and cost. To the best of my knowledge thisis the first time the Si5351A has been implemented in a product directly driving a QSD with two
outputs in quadrature (no divide-by-4 circuit).
The Clk0/1 Si5351A outputs are also routed to the transmit driver. During transmit, Clk0 and Clk1
are configured with a 180-degree phase difference, this allows them to cleanly drive the QDX RF
power amplifier, which is a Push-Pull configuration. It’s a very neat way to obtain the necessary
push-pull driver signals in a highly symmetric and low parts-count way.
Finally the Clk2 output is used as a separate signal generator, it is routed back to the QDXtransceiver’s RF port via the 100K resistor R19. This signal generator can be used by the QDX
diagnostic and self-performance measurement tools, to sweep the audio filter and sweep the RF
bandpass input filter. Clk2 is normally switched off, and when these functions are needed, it isenabled in conjunction with Clk0/Clk1 operating the receiver mixer.
An additional function of the 25MHz TCXO and its NAND gate buffer amplifier, is to provide theprecise 25MHz system clock for the STM32. The STM32 microcontroller contains its own internal
PLL clock generation systems that are highly configurable. In QDX, the 25MHz reference oscillator
is fed to the STM32 external clock input, and the STM32 is configured to provide:•

24.576 MHz clock for the 512 fs input to the 24-bit stereo I2S ADC chip (24.576 / 512 =
48,000) – fs is the number of clock pulses for each analog to digital conversion.

•

48 MHz clock for the STM32’s USB peripheral to enable it to provide USB functionality for
the USB sound card and the CAT Virtual COM serial port.

•

72 MHz for the main CPU clock

None of these signals are absolutely precise but they are very close to the target values and wellwithin specified allowable errors for the protocols involved (USB, I2S). The 72 MHz CPU clock is alittle below the maximum allowable 84 MHz for this processor, but this is acceptable since theperformance at 72 MHz is plenty adequate to achieve all desired functionality for QDX.

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3.4 Solid state transmit/receive switch
During transmit, the receiver must be
disconnected from the RF port of QDX
because a 5W signal is 45V peak-peak
which would be too much for the receiver
input mixer IC3.
A spectacularly high “Off” isolation is not
required, only sufficient attenuation toprotect the mixer. During transmit, the Digital
Signal Processing of the I & Q channels issuspended and there is no audio output of
the receiver. In the QDX application a totallyclean click-free transition between Receiverand Transmit states is not needed – awelcome relaxation compared to thedemands of a CW transceiver.
The transmit/receive switch is implemented
by a single BSS123 MOSFET. The source is
at DC ground (via 47 uH inductor L4). The
control signal from the microcontroller
switches the MOSFET on or off. Capacitor C35 close to the MOSFET gate is found to benecessary to prevent inductive pickup of the 5W RF from partially switching on the MOSFET.
During receive, the transmitter drivers are all at zero potential, so the transmitter power amplifiertransistors are in a high impedance state and do not affect the receive operation.
This simple transmit/receive switch very conveniently avoids the need for relays. Relays may begreat for some applications but if a relay was used here for transmit/receive switching it would addbulk, weight, cost, and reduce reliability.

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3.5 Switched receiver band pass filter
In any receiver, performance is improved by excluding strong out of band signals using a band
pass filter. In QDX a rather simple series resonant L-C filter is used, which although limited, still
provides some useful protection to the mixer. The mixer is a Quadrature Sampling Detector which
is a high dynamic range, high IP3 mixer and therefore has excellent intermodulation performance.
Together, the BPF and QSD provide a high performance receiver front end.
The band selection switch is a FST3253 dual 1:4 analog multiplexer IC, containing in other words,
two 4-way switches.
The switch connects one of Y0 to
Y3, to the W pin; and at the sametime one of X0 – X3 to the Z pin.
The selection of which of the four
possibilities is connected, ismade by two control lines named
A and B (pins 14 and 2
respectively) which areconnected to output pins of the
STM32 microcontroller.
The Y-W switch connects one of
capacitors C29, C31, C28 and
C30 (for 20, 30, 40 and 80m
respectively) to pin 9 (W).
The second switch X-Z thenselects one of four taps ofinductor L12.
Inductor L and capacitor C
values are chosen such that the
LC resonance is centered on thedigi modes section of each band.
The whole BPF is DC-biased to
2.5V midrail via the trifilar phasesplitting transformer T2 of the Receive mixer.
More sophisticated Band Pass Filters could be used but would come at the expense of complexity,
board area, and cost.
NOTE
The Fairchild FST3253 is no longer manufactured in the 16-pin SOIC package, it is only available
in the much smaller 16-pin TSSOP packaging. Texas Instruments produce an equivalent device,
the 74CBT3253C which is still produced in 16-pin SOIC packaging; this TI version is used on
current PCB revisions of the QDX. In this technical description, the old part number FST3253 is
used.

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3.6 Double-balanced Quadrature Sampling Detector

This circuit implements a double-balanced Quadrature Sampling Detector, mixing from RF to
baseband. The FST3253 is a dual 1:4 multiplexer which is often seen in QSD circuits. It has fastswitching times and very low on resistance of only a few ohms. The input signal is switched by thequadrature LO to each of the four integrating capacitors C39, C40, C41 and C43 in turn, for 90degrees of the RF cycle each. The result is that the audio difference (beat) between the RF input
and LO input appears across each of the four integrating capacitors, with four phases at 0, 90, 180
and 270 degrees.
The 33nF capacitors and the low source resistance results also provide a relatively fast roll-off ofthe audio response. This is effectively a narrow band pass filter since any incoming RF more thana few 10’s of kHz away from the LO frequency is greatly attenuated. The QSD is inherently a very
high performance mixer design with high third order intercept and dynamic range, and low loss
(0.9dB).
The FST3253 dual switch is often connected with the two switches simply paralleled together(which does halve the switch ON resistance). But I prefer the double-balanced mixer configuration
which provides higher performance. The double-balanced configuration requires two RF inputs
180-degrees out of phase (opposite to each other). This is provided here by trifilar-woundtransformer T2.
R17, R18 and C38 provide a 2.5V DC bias to the RF signal through the mixer and audio preamplifiers. It also provides mid-rail DC bias to the Band Pass Filter component switches in theprevious section. This simple bias does not source or sink any significant current due to thebalanced nature of the system, therefore no buffering is required.
Note that the Clk0 and Clk1 signals from the local oscillator have 47-ohm resistors in series; somepeople say this reduces ringing artifacts in the detector though I have not been able to find any
evidence for this.

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3.7 Low noise balanced
differential pre-amplifiers
The same LM4562 dual operational amplifier
chip is used in QDX as is used in the QCXseries CW transceivers and QRP Labs
receiver module. This is a low noise op-amp
providing an excellent combination of highperformance and low cost.
Low noise is important for the pre-amplifierbecause it determines the overall sensitivityof the receiver (in conjunction with losses of
the RF front end).
This part of the circuit was substantiallyredesigned for the Rev 3 QDX PCB, due tothe fact that the formerly used AK5386 ADC
chip had become obsolete. The chosen
replacement, PCM1804 claims even higher
performance than the AK5386 but the
PCM1804 has a differential input. One input
could be ignored, feeding the I and Q signals
into only one terminal of each ADC (left andright); however, this would sacrifice 6dB ofthe potential 112dB claimed dynamic range of
the PCM1804.
The circuit used here is derived from the
instrumentation amplifier topology that usually
has three op-amps; the third op-amp being aunity-gain difference amplifier of the outputsof the first two. An instrumentation amplifierarrangement brings several key benefits inthis application:•••

Excellent balance means common
mode noise signals are canceled out
Very high input impedance makes theloading on the Quadrature Sampling Detector capacitors very light, as well as balanced
Provides positive and negative phase I and Q signals suitable for feeding the differential
PCM1804 ADC chip

The disadvantage is that we require twice the number of op-amps and more resistors andcapacitors – but you know what they say about free lunches.
In the circuit shown above, the ADC chip itself, having differential input, performs the differencefunction so no 3rd op-amp is required in the instrumentation amplifier. The “Common” input wasused with a resistor network to provide DC bias to the instrumentation amplifier inputs. However it
was found that in circumstances the LM4562 op-amps could “lock up”, particularly IC7, with its
output near the supply rail. On Rev 4 PCBs the DC blocking capacitors were shorted out postproduction; on Rev 5 and above, the DC blocking capacitors and the bias network are removed
completely; the DC mid-rail bias is fed through all the way from T2.
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The 4.7nF capacitors provide high frequency attenuation beyond the 48kHz sampling bandwidth.
The 10nF capacitors C58 and C60 are recommended by Texas Instruments in the PCM1804
datasheet and who am I to argue.
The gain of the instrumentation amplifiers is only about 20dB and is chosen to best match the
dynamic range of the PCM1804 ADC in the application.
The I and Q channel contain all necessary phase and amplitude information to allow the following
SDR to demodulate any mode, although only single sideband is used in QDX.

3.8 Analog to Digital Converter

The PCM1804 Analog to Digital Converter (ADC) chip is a high performance stereo ADC chip withdifferential input. Its purpose is to convert the analog I and Q-channel signals to a digital
representation to allow digital signal processing by the microcontroller, which implements a
Software Defined Radio (SDR). The PCM1804 specifies a 112dB dynamic range and is used at 48
ksps (kilo samples per second).
All the Texas Instruments PCM1804 datasheet recommendations in regard to layout and supply
decoupling with tantalum capacitors, are followed (they most probably had good reasons forwriting all that stuff and even if they didn’t, I’m not about to take risks here).
In this design, the PCM1804 chip acts as Master of the I2S interface and the STM32 I2S
peripheral acts as Slave. The STM32 generates a 24.576 MHz clock signal (I2S_MCLK) which is
512 x the sample rate (i.e. 48,000 x 512 = 24,576,000). From this, the PCM1804 generates the bit
clock (I2S_SCLK) at 3.072 MHz (64 bits per sample – two 32-bit fields, for Left and Right
channels, of which 24-bits are populated). The processor also generates the Frame clock (LRCK)at 48 kHz. The I2S_RESET signal is supplied by the microcontroller and must be activated once at
power-up to reset the ADC.

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3.9 Embedded Software Defined RadioThis schematic fragment contains the whole microcontroller section around the STM32F401RB
microcontroller.

The microcontroller section of the schematic includes a 24C64 (8K x 8-bit) serial I2C EEPROM
chip. Unlike the ATmega328 processors used in many other QRP Labs kits, the STM32 used here
does not contain internal EEPROM so for storage of configuration parameters it is necessary toeither use Flash storage or EEPROM. The problem with Flash storage is a limited 10,000 writecycle specification which may be a little low. Therefore an external EEPROM chip is used.
The 4-pin programming header at the bottom right of the diagram is for factory use only to install
the QFU bootloader on the QDX board. An onboard diagnostic LED is also used only for factory
validation. LED1 is a front panel LED on the QDX which shows basic status information such aswhether or not the unit is transmitting or receiving, or in bootloader (firmware update) mode. A
transmit-enable output switches on the transmit driver to enable the RF power amplifier.
Other than that, all the magic goes on in the firmware, which is not open source.

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The Software Defined Radio receiver (SDR) implements digitally a superhet receiver having a
12kHz Intermediate Frequency. The reason for doing this rather than processing directly at
baseband as a direct conversion receiver, is that it eliminates any issues with power line hum –the harmonics of 50Hz or 60Hz (depending on your country) and other noise which exists around
0Hz.
The receive signal path processing contains the following stages implemented in the Digital Signal
Processing (DSP):•

Retrieve a block of 32 24-bit audio I and Q samples from the ADC chip via the I2S bus•“Mix” to 12 kHz Intermediate frequency, including a Low Pass Filter in the factor-of-four
decimation process; the reason for the choice of 12 kHz as the intermediate frequency isthat the mathematics becomes relatively easy, as it is exactly one quarter of the 48 kHz
sample rate.

•

A Hilbert Transform applies a 90-degree phase shift to one of the I and Q paths relative tothe other; this is the equivalent of the analog all-pass phase shift network implemented
using op-amps in the QCX CW transceiver.

•

Sum or subtract the two paths to produce either the Upper or Lower sideband; QDX iscapable of demodulating either sideband but digital operations normally use upper sideband (USB) which is therefore the default operating mode.

•

Apply a digital audio filter, which has a passband from 150Hz to 3.2kHz•

Interpolate back to 48 ksps audio•

Send the 48ksps audio samples to the PC over the USB Digital Audio link.

The performance of the receiver was tested using the internal signal generator to inject a signal
into the QDX input at a defined offset; Argo software was used on the PC to determine thereceived signal amplitude.
Two different measurement runs were performed, with quite some elapsed time between them (17months!). In both cases, there is no change to the firmware but the measurements were performedon different prototypes. There is no attempt at amplitude or phase compensation, which does notappear to be needed (the results are already excellent).
The vertical scale in dB has an arbitrary zero reference; the important point is the difference
between the upper sideband (blue line), and the unwanted lower sideband (red line).

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In the first example (shown below) dating from 06-Apr-2020, the worst case unwanted sideband
suppression is at -64dB relative to the wanted sideband, which occurs at about 2,200Hz audio.

The second measurement (September 2021) shown below, shows a worst case unwanted
sideband -60dB relative to the wanted sideband, again at about 2200Hz audio. The difference of
4dB between the two prototypes will have been due to component tolerances and build-stylevariations. In both cases, this is really good performance!
The second measurement was painstakingly made with much finer resolution (lower frequency
steps) than the earlier measurement, which shows more clearly the behaviour of the Digital Signal
Processing.

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3.10 Embedded 24-bit 48ksps stereo USB Sound card
QDX has its own embedded 24-bit 48ksps stereo USB sound card! But there is not a real physicallittle piece of hardware strapped inside, like the PCB from a sound card dongle or something.
Instead, the USB sound card is actually coded in firmware directly by the STM32 microcontroller,
the same single microcontroller that is also doing everything else, in the design.
This is an extremely important design feature of QDX.
A radio transceiver having an audio input has a number of disadvantages:•

An audio cable is one more cable to have to plug in between your radio and your PC.

•

There may be level mis-matches between the PC generating the audio signal, and theaudio input of the radio. The PC will need to be carefully set up to set the volume such thatthe radio produces the expected RF proper power level, but does so without distortion(clipping, intermodulation distortion, splatter, however you want to refer to it). With the builtin QDX sound card, no level adjustment is necessary; set the volume to the max and rest
assured that the QDX will never be overdriven (in fact it CANNOT be).

•Audio cables are a recipe for picking up noise, particularly in most of our lab environments
where we power our equipment from grid power. You will often see 60 Hz (US) or 50 Hz(most of the world ex-US) harmonic lines on spectrum waterfalls. Ground loops can be hardto eliminate. With QDX and its built-in sound card, numeric audio sample data is transferredback and forth between the PC and QDX with zero loss, zero distortion, no hum or othernoise. The quantization error due to sampling at 48ksps is minimal but is in any casematched by WSJT-X so a higher sample rate would not achieve an increase inperformance. The benefit of direct digital numeric transfer between PC and QDX, without
any conversion errors or noise, is enormous!

You can buy cheap USB dongles and connect them between your radio and your PC. But the
dynamic range IS limited, and not just by the 6dB/bit theoretical relationship between bit depth anddynamic range, but also by the actual noise floor of the converter (the least significant bits are inother words, just noise).
In contrast, QDX uses a 24-bit stereo ADC chip with 112dB dynamic range specification. If you
look at the specification of any expensive USB external sound card, and the price of getting
anything like 112dB, you will appreciate the true significance of this.
All Digital Signal Processing in QDX is carried out at full floating point resolution, not limited to 16bits as many SDR implementations do. In its communication with the PC this is rounded to 24-bits
and transferred via the USB Digital Audio device class. WSJT-X only uses 16-bits so the PC
sound system will automatically re-sample but still, the 24-bit USB sound is sent by QDX tosupport any future PC applications which use a higher resolution.
The result is a very high performance radio receiver, and equally, an excellent performancetransmit side due to the zero-loss, zero-noise transfer of audio digitally from the WSJT-X synthesis
straight to the QDX frequency analysis.

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3.11 CAT control serial interface
QDX includes an emulated virtual COM (serial) port, all in the same STM32 processor that does
everything else. The STM32 processor therefore actually includes THREE USB device classes:
1. Compound device class (a container for more than one other device classes)
2. Digital audio
3. Virtual COM (serial) port
The introduction of the compound device class makes QDX effectively, contain a virtual USB hub,to which is connected a Digital Audio USB device (the USB sound card) and a Virtual COM SerialPort. This allows a single USB cable to be used between QDX and the PC, the USB cabletransfers both digital audio and serial data seamlessly between the two.
An important benefit of the CAT control interface is that it allows QDX to have a flexible tunable
VFO and band switching, all without any actual physical controls. No physical controls are at allnecessary because everything is done under the control of the WSJT-X software as CAT
controller. Elimination of display and physical controls is not just a matter of lowering costs, it also
makes the QDX extremely easy to use. Want to change bands? Simply click the drop-down in
WSJT-X and QDX will automatically change bands. Want to use a different frequency (forexample, change mode to WSPR, or perhaps even change software application to JS8Call)? The
software application will automatically switch you to the correct frequency for the mode you select.
Another advantage of CAT control is that WSJT-X contains an option to effect the transmit/receive
switch either by VOX (Voice activation, or in other words, the presence of audio) or by an actual
CAT command to enable transmission. QDX also supports both methods, there is an entry in theconfiguration screen that allows VOX to be enabled or disabled. The default setting is “disabled”,
which means a CAT command is necessary to switch from Receive to Transmit (and back).
The danger of VOX is that it is all too easy to accidentally have the QDX selected as the soundcard for system noise outputs etc… then when a WhatsApp message from someone arrives, and
the PC does its fancy “pling” sound, well that’ll be faithfully converted to a series of audio tones by
QDX and transmitted accordingly. So disabling VOX and insisting on a proper CAT command toswitch on to transmit, is the recommended modus operandi.
QDX also has a rich user interface via the Virtual COM port, by connecting a terminal emulator on
a PC, to the virtual COM port. This is a whole blissful and enormous topic by itself, it allows
interaction with all the built-in analysis tools in QDX as well as the configuration screen andfeatures such as factory reset and firmware update.

3.12 Audio frequency analysis
Determination of the audio frequency arriving from the PC software (for example, WSJT-X) is key
to the whole operation.
My first thought was a Fast Fourier Transform (FFT). However that quickly turns out to be a lot
more difficult than it at first sounds. For the FT8 mode for example, the audio range is a 3kHz
band. It isn’t so trivial to set up a FFT with 6.25Hz buckets, that wide, and would require asignificant amount of processing power, such that I would need to use a more powerful (andexpensive!) microcontroller than I’d really like. Furthermore WSJT-X modes are heavily optimized
such that the period is the inverse of the tone spacing, so I might need to do TWO interleaved FFTanalysis runs offset by half the tone spacing. Then there’s the problem of the frequency slide when
QDX assembly Rev 1.18d

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a tone changes. Furthermore, the QDX would have to know which mode was being transmitted, inorder to configure the size (and hence speed) of the FFT buckets. I started to shudder and shake!
I soon realized that FFT would be the sledge hammer approach and there is a much simpler andfaster way, which also offers higher accuracy.
The simple approach is to detect the zero crossings of the sinewave, and carefully time the period
between zero crossings. Frequency is the reciprocal of Period, so once you have the period, amoment later you know the frequency too, and the problem is solved. It turned out to be very easy
to implement this method, and the accuracy is rather impressive. Experimentally, a SINGLE
CYCLE measurement of the frequency at 1500Hz audio frequency leads to frequency accuracy ofaround +/- 0.05 Hz. More than enough!
NOTE: 1500Hz is an unfair frequency to measure at, because each cycle fits into an exact numberof audio samples; 32 audio samples at 48,000 cycles per second make up one 1500Hz audio
cycle. This sets us up for an extremely low error cycle measurement. So to make things fair, I
actually offset the frequency for the test, to around 1520 Hz, so that the cycle doesn’t contain anexact number of samples.
In actual fact we don’t really need to update the frequency of the Si5351A Synth 1500 times per
second… that’s excessive. QDX has configuration options that allow you to specify an averaging
period, where a specified number of sound samples AND a specified number of audio cycles must
both be exceeded, in order to terminate the audio cycle measurement. The default parameters areset for a minimum of one cycle, but a minimum of 480 samples. 480 samples is 0.01 seconds andsince we never go as low as 100Hz, the dominant criteria is therefore a measurement of at least
0.01 seconds.
With this default configuration, we get 100 audio frequency measurements per second, andexperimentally the frequency accuracy is better than +/- 0.002 Hz. Amazing.
As for the details of the analysis, study the following hypothetical sinewave:

Remember that we have a built-in 24-bit 48ksps stereo USB sound card. Accordingly we get
48,000 samples per second. In this example I am assuming WSJT-X is used; WSJT-X outputs 16bit samples, so the extent of the sinewave data is from -32,768 to + 32,767. The sound samples
are received from the PC via the USB cable and are absolutely FREE of any noise. It’s a losslessand zero noise transfer between the WSJT-X software on the PC, and the QDX’s firmware. The

QDX assembly Rev 1.18d59

sequence of sample numbers received are EXACTLY what the WSJT-X software generates in its
audio software-DDS.
We also know that the sound samples arrive at exact equal intervals of one 48-thousandth of asecond. This is effectively the time base of our period measurement. It is not necessary to run any
kind of actual timer inside the microcontroller, because in this instance we already have the luxuryof knowing we get exactly 48,000 samples per second from the PC.
In the graph, the sine wave is shown starting at sample 0, with amplitude zero. In general, except
in special cases (such as 1500Hz as mentioned above), we won’t have a subsequent zerocrossing indicated by a sample with value zero. We will have a sample ABOVE zero, followed by asubsequent sample BELOW zero. It is easy to detect this situation in the firmware code, just bylooking for two successive samples, the first one being positive and the second one negative.
In QDX the firmware detects positive-to-negative zero crossings but it could equally well be doneon negative-to-positive zero crossings, it is of no importance. It would even be possible tomeasure the frequency of half-cycles, with half the accuracy. But this kind of extremism doesn’t
seem warranted, unless in future some kind of mode with very fast measurements is needed or
perhaps, we need to quickly measure the frequency of very low frequency audio signals.
Once we have a positive value followed by a negative value, and we know that they are 1/48000thof a second apart, we now want to find the exact time of the zero-crossing itself. This is also quite
easy with a straight-line (linear) approximation. For small angles (in radians) the well known
approximation holds: x = sin(x).
The period of the measurement is then given by the fractional times calculated by interpolation, atthe start and end of the sample, plus the number of complete sound samples that occur
BETWEEN the zero crossings. This is a “number of samples”, and we know that there are 48,000
samples per second, so now it is easy to calculate the time in actual seconds which is the Periodof the audio cycle; then the frequency is 1/Period, and we’re done.

An example
To give a worked example, consider again the above simulated sinewave samples (generated in a
spreadsheet). For ease of viewing, there are not many samples per cycle, in other words, this is amuch higher frequency than we will be dealing with in practice; but it serves to make the principles
relatively easy to understand.
We see audio sample #5, which happens to have a value of +8481 (according to the
spreadsheet). Now the next sample, audio sample #6, has a value of -10126. We want to know
the fractional interval (in samples), that is at the start of this sinewave cycle. I have called this time
t1 in the diagram. Time t1 (as counted in samples), is a fraction 10126 / (8481 + 10126). This is astraightforward linear interpolation between the values at sample #5 and sample #6. The result
calculates to be 0.544203794 samples.
The next zero crossing occurs between sample #16 (value +6813) and sample #17 (value 11743). Now we want to know the time t2, which is the fraction of a sample that occurs between
the time of sample #16 and the zero crossing. I hope that you can see that this fraction is given by,in this case, 6813 / (6813 + 11743) = 0.367158870.
Next we can see that there are 10 complete sample periods in the cycle, between our two
fractional pieces t1 and t2. The complete period, measured in audio samples, is therefore t1 + 10
+ t2, which is 0.544203794 + 10 + 0.367158870 = 10.911362664.

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By now, we have determined that our audio cycle being measured, has a period of 10.911362664
audio samples. We also know that there are 48,000 samples per second. Therefore the Period of
the cycle in seconds, is 10.911362664 / 48000 = 0.00022732.
The frequency of the audio tone is therefore 1 / 0.00022732 = 4399.08 Hz.
All without anything too complex, no complicated DSP, Fast Fourier Transform, etc. Love it!

Error considerations
Now like any good scientist, we should consider the possible sources of error in this measurement
technique and try to understand if it is good enough for our purposes.
Thinking about it, I could envisage three possible sources of error: audio amplitude error, timing
error, and interpolation error (the straight line linear interpolation). Let’s consider each of these inturn.

1) Audio amplitude error:
WSJT-X produces samples with 16-bit resolution. Other digimodes software may offer 24-bit
resolution but I have not come across any popular modes or software that do this. Neither do I
think there are any that provide BELOW 16-bit resolution. Since most users will use WJST-X, andother software anyway has similar facilities, I will use WSJT-X for the purposes of discussion.
A 16-bit resolution provides 65,536 amplitude values per sinewave cycle. That’s a theoretical (andpractically achieved, since we have the built-in USB soundcard and therefore a lossless, noiselesstransfer mechanism) 96dB dynamic range in the information transfer between WSJT-X and QDX,
a very accurate transfer indeed.
And yet – there are two scenarios where this can be dramatically reduced.

The first scenario is where the operator uses the output amplitude slider at the bottom right of the
WSJT-X screen, to something less than maximum. On a more traditional digital modes station
involving a SSB transceiver, this would be done in order to avoid over-driving the SSB
transceiver’s audio input, causing splatter. There is NO POINT to doing this on the QDX, since the
QDX RF output simply cannot be overdriven. The QDX RF output simply is, it is the perfection of a
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RF signal without any spurious residual carrier or attenuated unwanted sideband, without any
audio harmonics due to over-driving and clipping.
But perhaps the operator wants to reduce the transmission output power, and uses the amplitude
slider to do this? Again there is NO POINT, because QDX does not transmit at reduced output
power. It’s all or nothing. There is no in-between.
Therefore the operator should be advised for best accuracy, to leave the volume slider at the
maximum value, as there is no point to any lower value choice anyway. I found I was able to movethe slider to very low volumes anyway before the frequency measurement accuracy significantly
deteriorated. So this is not a particularly serious concern. But the recommendation still remains,just leave the slider at maximum.
The second scenario where lower resolution sinewaves can be generated by WSJT-X, is during
the key-down and key-up instants, where WSJT-X applies (I believe), a raised cosine amplitudeenvelope to the beginning and end of the transmission. For the sake of design simplicity andbecause it is much less critical in digital FSK modes, than on/off keyed modes such as CW,
amplitude envelope shaping has been omitted from the QDX design.
So there is no point to WSJT-X sending reduced amplitude sine waves at the beginning and endof the transmission key-down, since QDX cannot implement that anyway. Of course, WSJT-X
does not know this, and there is no way to turn it off, so it will happen anyway. We can’t “advise”
WSJT-X software in the same way as we could advise the operator to leave the slider at maximum
setting, in the previous discussed scenario.
At very low amplitudes, there would be a much more significant quantization error due to thelimited amplitude resolution, that has the potential to increase the frequency estimation error
substantially at very low amplitudes such as when the raised cosine envelope shape starts. This
could cause a frequency estimation with a significant amount of error, and the Si5351A
Synthesizer to start producing RF at an incorrect frequency.
To avoid this possibility completely, the QDX has a configuration (“Rise threshold”) which allows it
to ignore any audio cycles until a certain percentage of full amplitude has been reached. This canbe set to a high percentage value such as 80 or 90%. This will ensure that there is no initialfrequency value with a high error. Transmission will not occur until the 80% threshold (forexample) is reached.
As a side-benefit, this will also FORCE the operator to keep the volume slider at the maximum
value, or there will be no RF output! The “advice” will become binding.
The threshold should not be set to a very high value such as 99% because at high frequencies,
where the cycle is made up of a smaller number of sample values, there may be no sample value
near the actual peak of the sinewave, and so it may take several cycles for perchance, a value toexceed the threshold.
There is a similar “Fall threshold” setting to switch off the transmission at a defined amplitude
threshold, before the raised cosine trailing edge of the amplitude envelope generated by WSJT-X
takes the amplitude to a very small number and causes a potentially inaccurate frequency
measurement.
“Rise threshold” has a default value of 80% and “Fall threshold” has a default value of 60%; it isunlikely the operator would ever want or need to change these from the default values. The default
values are not critical but they should be neither very small, nor very large; and furthermore thefalling edge threshold should be somewhat smaller than the rising edge threshold, otherwise keyup will be effected prematurely.
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When the operator uses the maximum volume setting, and thresholds are implemented to avoid
the possibility of high calculation error at very low amplitudes which could occur during the raised
cosine leading and trailing edges, the effects of amplitude error are practically speaking, of noconsequence.

2) Timing error:
USB audio devices transfer data to and from their PC host at an agreed rate. In the case of QDX,
it appears to the PC as a 48ksps (48,000 samples per second) 24-bit stereo USB sound card.
However, in common with all such USB audio systems, there is no facility for synchronization
between the PC USB host, and the USB device. In other words, there is no 48kHz clock signal
between the two. In the Full Speed USB protocol, the host and device communicate via data
packets sent 1000 times every second.
Now it can get all rather complicated… because the host may send data to the device at notexactly 48kHz, and the device may send data to the host at not exactly 48kHz either. There are
specifications in the USB standards that define the worst allowable error percentage. Now bothends of the connection have to try to deal with the fact that there is ALWAYS going to be adifference between them. Even if both 48kHz crystals derive their sample rates from a quartz
crystal reference, the crystals will not be at exactly the same frequency.
In the case of QDX, there is a 25MHz TCXO that operates as the CPU clock input. All signals arederived from this clock, including the 72.something CPU clock, and the 48MHz USB peripheral
clock (well, CLOSE enough to 48MHz), and the I2S peripheral master clock which is at 512 times
the sample rate (each sample has 512 clocks); or 48kHz x 512 = 24.576 MHz.
This 24.576 MHz signal is derived internally in the QDX, by a PLL multiplication factor of 117, thendivision factors of 17 and 7. So 25 * 117 / (7 * 17) = 24.579832 MHz. Dividing by 512 gives anaudio sample rate of 48007.5 samples per second! As well as this synthesis accuracy limitation
there’s also the smaller imprecision in the 25MHz TCXO oscillation value.
Luckily, this small 7.5 samples per second difference is well within the allowable percentage error
in the USB sound device specification. On the PC side, the built-in USB driver has to deal with thisdifference, in addition to the fact that its own master clock will also not be truly at precisely 48kHz;
it does so, I believe, not by digitally re-sampling to what it thinks is exactly 48ksps, but by somekind of software phase lock mechanism.
On the incoming data from the PC over the USB interface, we’re also not going to be getting data
samples at exactly what WE believe is 48ksps (our belief also being slightly inaccurate, itself). So
what do we do… certainly I’m not going to dabble in software PLL clock recovery; an option could
be do occasionally drop a sample if there are too many, or to duplicate a sample from time to time
when there are, in my judgement, too few.
But in the end, after all this waffle, none of it is necessary. It is perfectly acceptable to do
NOTHING about the lack of synchronization, just nothing at all! It is fine to pretend that we getexactly 48,000 samples per second, and analyze them all perfectly as though everything wasexact. If there is any sample rate error, it will simply produce in the calculation result, a
proportionate error in calculated frequency. Even a 1Hz error, if it happened (and frankly, I havenever seen even a 0.01Hz error), would be of no consequence to operation of the transceiver,alongside much more significant errors such as the precision of the RF synthesis.
In the end therefore, the conclusion is that timing errors are of no consequence.

QDX assembly Rev 1.18d

633) Interpolation error
Remember the approximation we have used for the zero-crossing interpolation that determines theexact moment of the zero-crossing event? Mathematically, the approximation is that for smallvalues of x, then sin(x) equals x. Put another, less mathematical way, we can draw a straight line
between the two points we want to interpolate, which makes the calculation easy.
What can cause this to generate error? Well, when we are no longer dealing with “small values ofx”. Practically speaking in our case, this occurs when there are a lower number of samples peraudio cycle.
Consider an extreme example, for the sake of demonstration – here is a an 8.4kHz sinewave(much higher than we will ever be dealing with):

You can see that the sample points, indicated by the X’s, are quite far apart. The zero-crossing
time, or the time between the two samples adjacent to the zero-crossing, is a relatively large
fraction of the cycle period. Our “x” is therefore not so small as it was before. The straight linesthat previously didn’t look at all bad, now look a lot less like the sinewaves that are being
approximated.
Now consider this sinewave which happens to be about 1.27 kHz:QDX assembly Rev 1.18d

64

Obviously it LOOKS a lot better but we can also see that using straight lines at the zero crossing,
will be a very much better approximation to the sinewave, than the high frequency case.
We can therefore say that when the sample rate is fixed at 48ksps, a low frequency audio
sinewave is a lot more accurately interpolated, than a higher frequency audio sinewave. In otherwords, the errors get bigger, and quickly so, as we increase the audio frequency. This can alsoeasily be seen experimentally by generating a signal in WSJT-X and then observing the measured
frequencies (this can be done using QDX’s own built-in tools, as described elsewhere in thismanual).
The error in frequency measurement is also reduced by measuring over more than one cycle.
Since we are adding a fraction of a sample at the beginning, and a fractional of a sample at theend of each cycle, and counting the number of samples in between – any error on the fractional
estimations (due to linear interpolation error) will be reduced by counting multiple samples. For
example, if we measure for 2 cycles, the error will be halved. If we measure for 10, the error will be
reduced by a factor of 10.
As mentioned earlier, it is quite unnecessary to count the frequency so often as a SINGLE cycle. If
we are measuring say, an audio signal of around 3kHz – do we really need to measure it 3000
times per second? Not practically speaking, no.
So the QDX has a configuration built in, which specifies a number of cycles that must be
measured, and a number of samples that must be measured. BOTH of these parameters must be
satisfied, before the measurement activity completes. The default values are 1 cycle, and 480
samples. 480 samples is a time duration of 0.01 seconds. In this configuration, a very low
frequency signal, below 100Hz (0.01 seconds period) would be measured properly. And at any
higher frequencies than 100Hz, the minimum samples criteria will dominate, and ensure that the
averaging period is automatically increased for higher frequencies! For example at 3kHz, 480
samples will contain 15 cycles, so the error in the fractional period estimation due to interpolation
inaccuracy, will be reduced by a factor of 15. In this way, high frequencies do more averaging thanlow frequencies; we get 100 frequency measurements per second.
Again these configuration parameters are not particularly critical, and again, the default values will
suffice for all imaginable purposes and most likely the QDX operator will never want or need tochange these, other than perhaps for his own curiosity of investigation.

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Sinewave interpolation
As an aside – for interest only – I did investigate the possibility of more advanced interpolation
methods than linear (straight line). I spent some enjoyable time experimenting with a method
which measures the amplitude of the sinewave (looking at the incoming samples) and then fits thetwo samples either side of the zero crossing to an actual sinewave rather than a straight line.
It did provide some improvement, some observable reduction in error. However, the error
reduction was not as great as I had expected, and I suspect this is due to limitations in the floating
point accuracy of the arc-sin implementation, perhaps (speculation) a limitation to the number ofterms of a polynomial expansion? Furthermore, although the STM32 processor used has a floating
point unit, the calculations were still too slow at the 72MHz CPU clock speed used. In
experimental measurements the calculations were too slow to be able to perform single cycle
frequency estimation at the upper end of the frequency range, which is exactly where we mostneed the improved accuracy; multiple cycles were therefore necessary and it was found that the
improvement in accuracy of averaging simple straight line interpolation over several cycles, wasmore than the improvement in accuracy of the slow sinewave interpolation!
Furthermore other inaccuracies could creep in, such as the estimation of the maximum amplitude;
or we’d have to assume the maximum amplitude sinewave, which would work less well on therising and falling keying edges.
I toyed briefly (very) with the idea of trying to implement a three (or more) point sinewave fit, but
dismissed it as it would involve even more calculation and be even slower. Polynomial fits wereanother possibility considered.
But frankly – in the end – the simple linear interpolation is very reliable at the frequency range ofinterest, and is already SO good, that the idea of trying to pursue more advanced approaches,
seemed, particularly after a day’s worth of investigation, to constitute an unnecessarily frivolous
indulgence in mathematical decadence.

Errors conclusion
The frequency estimation by cycle timing method employed in QDX is exceptionally accurate, farmore precise than needed for any digi modes likely to employed on HF bands during QDX
operation (I cannot imagine any exceptions). There are a few boundary cases where accuracy
could conceivably be impaired, and though this is actually still unlikely to be of any practicalsignificance, configuration parameters are provided to mitigate and control the errors, with suitable
default values that will be excellent for all conceivable use cases.
As a final word on audio frequency estimation, I want to also mention that WSJT-X is said totransition evenly between two frequency symbol tones, by implementing a raised cosine fallingenvelope shaping on the finishing tone, at the same time as a raised cosine envelope shape on
the newly starting tone. As discussed previously, QDX just doesn’t do envelope shaping. However,I believe that to all intents and purposes, that this fancy raised cosine tone envelope shaping isequivalent to a gentle “frequency slide” between the old and new tone frequencies. Perhaps it iseven exactly mathematically equivalent, I don’t know.
At any rate – for any mode WSJT-X generates, QDX will continuously measure the frequency andupdate the transmitter output frequency accordingly; at 100 updates per second (the default
configuration) this is plenty sufficient to achieve smooth transitions between tones; additionally thefrequency synthesis method used in the Si5351A itself causes a “slide” between frequencies not asudden discontinuous transition, because the PLL takes a finite time to settle on the new value.
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QDX contains its own test and analysis tools built-in; elsewhere in this manual is a description of
how you can observe these frequency transmissions yourself, using the built-in tools.

Si5351A fine control
Historically QRP Labs products have set up the Si5351A with an integer frequency, except for the
Ultimate3S QRSS/WSPR transmitter which has an algorithm during WSPR operation to optimize
the frequency steps for very accurate 1.46 Hz tone spacing.
The problem with QDX is that the QDX must faithfully transmit any frequency given to it. The tone
spacing may be very fine, or it may be coarse. QDX does not know. QDX knows nothing of thetransmission mode. All QDX does is receive audio tones from a PC and effectively modulate themto RF. The tones may be 1.46 Hz spaced in the case of WSPR, or 6.25 Hz in the case of FT8, or
many other possibilities. Therefore a fixed algorithm for WSPR such as Ultimate3S has, or an
integer frequency configuration such as used in the QCX CW transceiver, will not be optimum
here.
As an example, suppose we want an output frequency of 14,097,002.123 Hz and the firmware has
to decide the parameters to configure the Si5351A chip. So let’s say we chose a MS Synth dividerof 62 (abiding obediently by the even integer MS Synth divider rule). Now that means the internal
VCO must operate at 870,014,131.626 Hz. The multiplication factor from 25 MHz is therefore
870,014,131.626 / 25,000,000 = 34.960565265039996. Now that has to be represented as a + b /c where all are integers, b is in the range 0 to 1,048,575 and c is in the range 1 to 1,048,575 (band c are 20-bit numbers – so the max value is 2 ^ 20 – 1).
The question therefore is the choice of b and c to get the fraction 0.960565265039996.
Conventionally one may simply set c to 1,048,575 and set b to 0.960565265039996 x 1,048,575 =
1,007,224. Now we can calculate the output frequency:
25,000,000 x (34 + 1,007,224 / 1,048,575) / 62 = 14,097,001.845 Hz
The error in the output frequency is 14,097,002.123 – 14,097,001.845 = 0.228 Hz. It’s rather larger
than we might like; remember also that the error will be different every time and could be better orworse than this.
Well it turns out we took the cowardly way out by choosing the denominator (c) to be 2 ^ 20 – 1(1,048,575) and that there are better choices which will bring the ratio b / c very much closer to thedesired fraction 0.960565265039996. In fact the choice b = 416,333 and c = 433,425 works very
well… look, now the output frequency is:
25,000,000 x (34 + 416,333 / 433,425) / 62 = 14,097,002.123 Hz
The error, to three decimal places, is now 0.000. Bingo!
However the question is STILL, how to choose b and c to get this magical and wonderful result.
The answer is “best rational approximation” which delves into an area of mathematics called“continuing fractions”. Rather than spoil (and lengthen) this manual with my own garbled
interpretation, allow me to just refer you to Google and Wikipedia
https://en.wikipedia.org/wiki/Continued_fraction .
The implementation in QDX is a simple and fast iterative algorithm based on the information found
here, which rapidly converges on values for b and c which provide a very optimal ratio.

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3.13 Class-D Push-pull Power amplifier

An interesting power amplifier circuit was designed specially for this transceiver. There are four
BS170 transistors. Two are connected in parallel for the “push” side, and two connected in parallel
for the “pull” side. Each side is driven by two AND gates of a 74ACT08 quad AND-gate IC, inparallel to ensure strong and fast drive.
The AND gate drivers are clocked from two outputs of the Si5351A Synthesizer which arearranged to have 180-degree phase shift. This provides the necessary phasing for the “push” and“pull” sides of the amplifier, without requiring an input transformer. All the 74AC08 AND gates areenabled by a “transmit” or in other words, a “key down” signal from the microcontroller; only when
this signal is high, are the oscillator signals passed on to the MOSFET “gate” connections – this is
the operation of the logical AND function.
The outputs of the “Push” and “Pull” sides are combined by transformer T1.
This power amplifier design provides a number of useful features:•Class-D operation is efficient compared to Class C, or even worse, Class A/B linear which
as discussed previously, is overkill for this application. Class-D is not as efficient as Class-E(as used in the QCX CW transceiver), but it better suited to a multi-band transceiver, sincea Class-E amplifier has a resonant load circuit and this must be switched by band.

•

Push-pull operation provides 5 W output power at a lower voltage than would be needed ina single-ended amplifier (9 V supply produces about 5 W output).

•Push-pull operation cancels even harmonics which are therefore at a very low level (dueonly to imperfect cancellation due to slight differences in transistor characteristics).
Importantly the 2nd harmonic level is very low, which greatly simplifies the demands on the
Low Pass Filters that will subsequently clean up the signal after the amplifier.

•Push-pull operation allows us to use four BS170 transistors to spread the load, but they are
only connected in pairs so the driver only has to drive two paralleled gate capacitances not
four.

QDX assembly Rev 1.18d

68•Using an antiphase drive (the precise 180-degree out of phase signals from the Si5351A)
eliminates the need for an input transformer that is normally required in push-pull power
amplifiers.

•

All components are low cost and easy to replace in the unlikely event of failure.

As an example spectrum analyzer output, consider the following trace showing an 80m QDX. For
this measurement, a 50-ohm dummy load was connected directly at the output of transformer T1,
with no Low Pass Filtering.
The second harmonic is seen, about -70 dBc (70 dB below the fundamental) due to thecancellation effect of the Push-Pull amplifier. The subsequent even harmonics are even lower
amplitude. Only the odd harmonics are problematic and require filtering.

3.14 Switched output Low Pass Filters
Every radio transmitter always requires low pass filters at its output to attenuate any out of bandspurious emissions to sufficient level to comply with regulatory requirements (for example FCC in
United States).
In QDX the Low Pass Filters are carefully optimized to provide excellent performance with low
parts count. This can be achieved because the Class-D push-pull power amplifier design produces
a very low level of even harmonics.
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It is very important to reiterate that Low Pass Filter and Power Amplifier design must be
considered carefully, at an overall SYSTEM LEVEL. In fact this applies to the entire transceiver
design, but nowhere more importantly than the transmitter output stages, where the performanceis critical to your regulatory compliance. The requirements on Low Pass Filtering are determinedby the level of spurious emissions produced by the final Power Amplifier as well as levels in earlier
stages feeding into the power amplifier. One cannot simply transplant a circuit block from one
transceiver design to another, and expect the performance to be satisfactory. System level design
is critical.
In QDX there are three Low Pass
Filters. Each is made up of a 5element filter, with a 6th element asresonant trap at the 3rd harmonic.
The three filters are:
•

80m

•

40m

•

30/20mThe intrinsically low levels of 2nd
harmonic is what lets us share thethird filter between 30m and 20m.
There is a PIN diode switch either side of each Low Pass Filter. So at any one moment, one pairof switches is activated, to switch the appropriate Low Pass Filter into the circuit.
The discussion of this block of the circuit divides naturally into a two parts: the LPF design, and the
PIN diode switches design.

Low Pass Filters
The Low Pass Filter design was done using the Elsie program by Tonne Software. It is freewareand can be downloaded from here: http://tonnesoftware.com/elsie.html

80m:
For 80m this is the final schematic, having a resonant trap at the third harmonic:

The simulation shows:

QDX assembly Rev 1.18d

70Fundamental:
3.5 MHz
Second harmonic: 7.0 MHz
Third harmonic:
10.5 MHz-0.17 dB
-26.4 dB
-67.5 dB40m:
For 40m this is the final schematic, having a resonant trap at the third harmonic:

The simulation shows:

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Fundamental:
7 MHz
Second harmonic: 14 MHz
Third harmonic:
21 MHz-0.17 dB
-28.8 dB
-72.1 dB

30/20m:For 30/20m this is the final schematic, having a resonant trap at the 20m second harmonic,
28MHz. This is a departure from the 80 and 40m designs, which have the trap at the third
harmonic. The reason for this is that the 20m second harmonic is close to the 30m third harmonic.
This was found to represent an optimal compromise between 30m and 20m performance.

The simulation shows:

20m:
Fundamental:
14 MHzSecond harmonic: 28 MHz-0.12 dB
-73.8 dB

On 30m, while the second harmonic (20.2 MHz) attenuation is only about 12dB, the evenharmonics are in any case at a very low level; the more important harmonic is the 3 rd harmonic andfor 30.3 MHz the attenuation is about -45 dB which is very satisfactory.

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Low Pass Filter switching
The schematic of the Low Pass Filters and switching arrangements is shown below.

Very often relays are using for Low Pass Filter switching. However relays have several
disadvantages:• Bulky• Noisy• Expensive• Potentially unreliable (mechanical moving parts)• Poor isolation at RF, can be important in some applications (unless expensive relays used)

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Cost and bulk are of critical importance here. A better solution comes in the form of diodeswitching. Not just any old diode, PIN diodes. This is a fascinating topic and I did a great deal ofresearch and experiment to perfect the circuits used in the 50W PA kit http://qrp-labs.com/50wpa
and many of those ideas are carried over to the QDX design.
Ordinary diodes cannot be used for switching
because they could act as Rectifiers, switching on
and off in time with the RF cycle. A PIN diode,however, is different. It is a diode with a wide
undoped “intrinsic semiconductor” region between
the usual P-type and N-type doped semiconductor layers of a regular diode (Wikipedia image).
The important point is that this undoped “intrinsic layer” acts like a storage reservoir. When
incoming forward-biased current arrives, this “intrinsic layer” fills up with electrons. When the diodeis reverse biased, it takes time for all these electrons to empty out of the intrinsic layer. At highenough frequencies, the time available is too short! So the diode never “turns off”. It remains
conducting through the whole RF cycle.
PIN diodes behave like variable resistances. Their resistance is determined by the amount of DC
forward bias current. They are near-perfect variable resistors which add very little distortion to thesignal. In our case, we don’t care about variable resistors, all we want is either a very low
resistance (On), or a very high resistance (Off).
A problem is that PIN diodes tend to be expensive, harder to finditems. However, a solution comes along, the 1N4007 rectifier
diode! This is the big brother of the 1N400x series of rectifier
diodes, and has a reverse voltage rating of 1,000 V. By good
fortune, its internal construction in order to achieve this very high
PIV rating, is very similar to a “real” PIN diode. Hence the
1N4007 is often referred to as the “Poor man’s PIN diode”!
No doubt the 1N4007’s characteristics are not repeatable enough, or its resistance not linearly
controllable enough, or some such other imperfection, that it is not suitable for all applications. But
in our application where all we need is an On/Off switch at HF, a 1N4007 costing mere centsfunctions just as well as a “real” PIN diode costing 50x the price!
1N4007 functions well enough as a PIN diode switch across the whole of HF to 10m band, andeven down to 500kHz.
What about dynamic range and third order intercept point IP3? A properly handled 1N4007 should
not impact the dynamic range or IP3. An excellent set of measurements by Claudio IN3OTD see
https://www.qsl.net/in3otd/electronics/PIN_diodes/PIN_diodes.html and https://www.qsl.net/in3otd/electronics/PIN_diodes/1N4007.html confirms the excellent characteristics of the 1N4007, which at
HF in this switching application are similar to other PIN diodes. Other experimenters havedetermined similar results. Broad conclusions are that a properly biased 1N4007 at HF (1.8 MHz30 MHz) has the following characteristics:•••“On” insertion loss less than 0.1 dB for 10 mA forward bias current“Off” isolation at least 30 dB (at worst case, 30 MHz frequency)
IP3 > +50 dBm

Therefore the use of 1N4007 PIN diode switches should result in very little loss of transmitter
output power, very little loss of sensitivity on receive, and will not degrade the excellent IP3 (anddynamic range) characteristics of the QDX transceiver with its high performance double balanced
Quadrature Sampling Detector front end.
QDX assembly Rev 1.18d

74What of “proper biasing of PIN diodes”?
In my opinion the most excellent reference for the use of PIN diode switches for HF
Transmit/Receive is the work of Don W6JL. Don has a very informative page on QRZ.com https://
www.qrz.com/lookup/w6jl and a long article “Homebrew PIN diode QRO QSK system” inFunkamateur, March 2016 which is available as a PDF on the internet see
https://www.funkamateur.de/tl_files/downloads/hefte/2017/w6jl_improved_qsk_system_mar_2016.
pdf. Study of W6JL’s work should be the first task of anyone wishing to experiment with HF PIN
diode switching. Don has a separate homebrew transmitter and receiver. The transmitter is 600W
and is permanently connected to the antenna and the receiver, via his solid state PIN diodeswitching system. Sidetone is via the receiver monitoring the transmitter directly! The switch
provides 137dB of isolation in the “Off” state. Amazing! We don’t need anything so extreme (orcomplex) here – but the principles are well described by W6JL and his work is inspiring.
It is possible to distill the whole topic into two important rules:••

To switch the PIN diode Off, apply a large reverse bias VOLTAGE. The voltage should behigher than the peak-to-peak RF being switched.
To switch the PIN diode On, apply a forward bias CURRENT; the more current, the lower
the insertion loss. At 10 mA forward bias, the insertion loss is only about 0.1dB. One canincrease it a bit further, to be on the safe side.

When the two “rules” are followed carefully, the performance of the PIN diode switch is excellent;
furthermore it is low cost and reliable to implement using inexpensive common components likethe 1N4007. In the QDX transceiver the switching task is considerably simplified compared to the
50W PA kit, because here there is no need to support fast QSK (full break-in) operation or provide
a receive bypass path.
To understand the operation of the switching, consider just ONE of the Low Pass Filter circuits, the30/20m circuit. At any time, one set of the three Low Pass Filter switch pairs are ON, and the othertwo are OFF.
Let’s consider first, the switch in the ON state. The partial schematic is:

QDX assembly Rev 1.18d75Remember that for a diode to be ON, it should be passing at least 10 mA current. In QDX, theforward current is set at 38 mA during transmit, this is determined by the bias circuit around Q1and Q2 which will be discussed in a moment. Why so high, you may wonder? The reason for 38
mA is that the diode, if biased at lower forward currents, increases the level of even orderharmonics due to its own distortion. 38 mA switches it ON so well that the distortion level is verylow.
Remember also that capacitors block DC, but allow the passage of RF; whereas inductors block
RF, but allow the passage of a bias current (Direct Current, DC).
In the ON state, the microcontroller I/O pin controlling the Low Pass Filter is set to highimpedance, resistor R12 (10K) to +5V then pulls the potential at the gate of MOSFET Q5 to +5V,
switching ON the Q5 MOSFET. The bias circuit can now drive 38 mA through each of L1 and L11,
through the respective diodes D3 and D6, through L4 and L10 respectively, the sum of thecurrents (76 mA) through L9 and through the Q5 transistor to ground. These flows are indicated by
the pink arrows on the diagram. The various capacitors block DC, only permitting the bias current
to flow through the components indicated by the pink arrows.
Now that a sufficient forward bias current flows in D3 and D6, RF will see a low resistance all theway from the PA, through C34 and D6 to the Low Pass Filter; and on the other side of the filter, RF
will pass easily through D3 and C2 right to the antenna. L1 and L11 block RF from escaping into
the bias current circuits.
In the ON state, we think about current (forward bias current). In the OFF state, we must think
about voltage (reverse bias voltage). Here’s the same schematic, labeled for the OFF state.

Remember the first rule of PIN diodes? “To switch a PIN diode off, apply a reverse voltage that islarger than the peak-to-peak voltage of the applied RF”. Now, 5W of RF is 45V peak-to-peak(assuming 50-ohms impedance) and therefore we need more than this, as reverse bias, to switch
off the diodes. Obviously this is not a voltage that is available from any supply rail! While Don
W6JL had available 350V from the screen grid of the valves (tubes) in his transmitter – we haveno such luxuries available here.

QDX assembly Rev 1.18d

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The very neat solution is to “borrow” a little energy from the PA output itself! Then voltage double
and rectify it. The result is a nice clean DC which is always more than the peak-peak of the signal,and plenty enough to reverse bias the diodes so they are always perfectly and completely OFF.
This doubler section of the schematic is formed by D9, D8, C32 and C27. The resulting voltage,
we shall call B+ in honour of valve (tube) circuits of old. Or HT if you prefer. It’s exact value will
depend on the impedances and other factors but don’t worry, it is sufficient for the proper diodeOFF condition.
Now this voltage passes through resistor R9, through inductor L9, and through inductors L4 andL10, right to the cathodes of diodes D3 and D6 respectively. The inductors don’t impede DC;
resistor R9 would impede DC but the current involved is near zero so practically speaking there isno significant voltage drop across R9. The anodes of diodes D3 and D6 are at a potential of about+0.7 V. Why? Because whilst this Low Pass Filter is OFF, one of the other two LPFs is switched
ON, and 38 mA current is flowing through each of its diodes, resulting in a voltage across thediodes of about 0.7 V.
So with B+ on the cathode and 0.7 V on the anode, the diodes see a large reverse bias voltages
and satisfy the conditions for the diodes to be OFF. They present a high resistance to the RF,
which cannot then flow through this Low Pass Filter. It is OFF.
The next thing to worry about is what happens when we are NOT transmitting. The signal during
receive, still needs to pass through the proper Low Pass Filter for the band of operation. Its
switches still need to be ON, and the other two LPFs OFF. Now there is no large 5 W (45 V peakpeak) signal to rectify and double to make a large B+ voltage. On the other hand, we also nolonger need such a large B+ voltage, because now the Low Pass Filters (and their switches) areonly seeing the small signals picked up by the antenna. To provide some voltage to reverse bias
the PIN diodes, D7 connects +12V (the QDX supply voltage) to B+ during receive, when the
doubler is not producing rectified RF (during transmit).
The other minor detail to consider, is that during Receive we aren’t handling anywhere near such alarge amplitude signal so we don’t really need such a large current as 76 mA (total for two diodes)
forward bias through the ON switches. A smaller current will suffice and will reduce the overall
QDX receive current, making it more battery friendly.
Here’s the bias
circuit used
BEFORE PCB Rev 4,that generates theappropriate forward
bias currents.
When in receivemode, the TX signal
from themicrocontroller is at
0V; this keeps Q2
MOSFET switchedoff.

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Q1 is a P-channel MOSFET and its gate is pulled to +12V when Q2 is off via 10K resistor R3,
meaning that Q1 is also switched off.
In this configuration current flows from the Vcc supply rail (+5 V) through the 1 K resistors R1 andR6, and the inductors L1 and L11, to the respective two diodes. The current in each diode is given
by Ohm’s law I = V / R. The voltage across the resistor is 4.3 V because the voltage across thediode is 0.7V approximately. Therefore the current flow in each leg is 4.3 V / 1000 ohms = 0.0043
A, which is 4.3mA.
So in receive mode, the forward bias in each of two diodes in the ON Low Pass Filter, is 4.3mA for
a total switch current of 8.6mA, which is not unreasonable.
When the TX signal from the microcontroller goes high, transistor Q2 switches on, pulling the gateof Q1 to near ground potential, thereby switching Q1 on. Now 12 V flows through Q1, and througheach of R2 and R4, and on to the diodes. The current contribution through each resistor is given
by Ohm’s law I = V / R. The voltage across the resistor is approximately 11.3 V because thevoltage across the diode is 0.7V approximately. Therefore the current flow in each leg is 11.3 V /330 ohms = 0.0342 A, which is 34.2 mA. This sums with the current from the +5 V supply rail, togive a total of 34.2 mA + 4.3 mA = 38.5 mA.
So in summary, due to this biasing circuit:••

In Receive, total switch current is 8.6 mA approximately
In Transmit, total switch current is 77 mA approximately

A new bias circuit for PCB Rev 4
The above mentioned transmit bias circuit worked however, it does suffer a problem: the power
dissipation in each of the 330-ohm resistors (R2 and R4), when using a 12V supply, will be (in asimplified analysis) 0.038 A * 11.3V = 429mW. That’s a large power dissipation in such a small(0603-size) SMD resistor, certainly exceeding its rating. The total effect of 858mW of additional
power dissipation during transmit, is to cause significant temperature rise in that area of the board– under the output transformer – which may lead to an incorrect conclusion that the output
transformer is running inefficiently and overheating.
A much better solution would be to use lower value resistors, and a much smaller voltage drop.
For example, smaller resistors switched to the +5V rail, rather than 330-ohms to the +12V rail.
That would work and resolve the resistor-out-of-spec problem, however really only moves thepower dissipation from the resistors to the 78M05 voltage regulator. To really make the system
more efficient would require a switched mode voltage regulator.
However, a switching regulator risks introducing noise into the sensitive receiver. To avoid this, thedecision was taken in the Rev 4 PCB to use a 1.5V bias, using a switched mode regulator that is
enabled only during Transmit. It therefore cannot interfere with the sensitive receiver.
A review of various commonly available inexpensive Switched mode regulator ICs wasundertaken. A common characteristic is that the efficiency drops quite considerably when:
1) Low current loads are used (for example, below 100mA)2) The voltage drop is a high ratio, such as 12V to 1.5V will be in this case.
Below are the efficiency curves for several common IC’s reviewed. All of them exhibit this quite
significant decline in efficiency.

QDX assembly Rev 1.18d

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Top left: TPS562201
Top right: TD1410
Bottom left: MP2315
Bottom right: TPS562208
At first sight it’s not terribly encouraging, due to efficiencies around 70%. However, when you
examine the power savings it is possible to see that there will still be a massive saving in power
dissipation; it is not necessary to get over 90% efficiency in order to achieve very substantial
improvements in the system.
Given the current semiconductor shortage, which makes it quite unpredictable to obtain IC’s at all,let alone at reasonable prices, and for the sake of the educational possibilities, I thought it would
be fun to design a discrete component buck converter. I had also been inspired, some time ago,by coming across this well designed discrete 3.3V buck converter as shown at this link:
https://hw-by-design.blogspot.com/2018/08/vrm-discrete-33v-buck-converter.html
I used the same concept for the pseudo push-pull switch transistor drive, but replaced the control
loop with a firmware one. The STM32F401 as used in the QDX has plenty of spare unused I/Opins, one of which can be used to generate a PWM output, whose duty cycle is varied to achieve

QDX assembly Rev 1.18d79

the desired output voltage; and one more as an Analog input to measure the output voltage andimplement the control loop to regulate it at the target voltage.
The parts count to implement the buck converter circuit therefore becomes quite low. Using the
microcontroller to form the control loop also has additional advantages; the SMPS can be easily
shut down during receive operation when high current does not need to be pushed through the
PIN switches; and the microcontroller is also able to choose the desired output voltage according
to user configuration.
The actual circuit used is shown
here (right).
Q1 and Q2 were already part of thebias current switching circuit in the
Rev 1-3 boards so the only
additional components are the NPNtransistor Q14, the schottky diodeD11, inductor L17 and several
resistors and capacitors.
1.5V is shown here as the output
voltage but this is under control ofthe microcontroller; firmware version
1_05 and above must be used, andthe voltage is set to achieve a bias
current that is specified in theconfiguration screens (defaulting to30mA).
Q1, D11, L17 and C49 form a quite
conventional high-side switch layout common in buck converters. The more interesting aspect tothis is the “pseudo push-pull” gate drive. I had to think about it for some time, to understand how
this works.
When the PWM signal from the microcontroller goes high, Q2 is switched on, and this pulls thegate of Q1 low, switching on Q1 very quickly. This is the “pull”, which happens when the P-channel
MOSFET Q1 (BSS84) is switched ON.
If that is as far as we took things, the buck converter would work, but it would have poor efficiency.
The reason for this is that there is an effective gate-source capacitance which holds a gate charge
on the MOSFET. When the Q2 switch is opened, removing the “pull”, the gate charge would
dissipate relatively slowly through R3. The problem with a slow transition of the switch is that it isan opportunity for power loss. Ideally we want the switch to be saturated ON, with zero resistance.
Zero switch resistance means zero voltage across the switch, and P = I x V with V being zerowould mean no power loss. When the switch is saturated OFF, with ideally infinite resistance, thencurrent would be zero and again, no power loss. The FASTER the switch transitions between the
ON and OFF states, the less time will be spent wasting power in the non-ideal intermediate phase.
This is why the slow switch-off wastes power, reducing conversion efficiency.
Adding Q14 solves this problem, it switches on and rapidly discharges the gate charge, causing
Q1 to be switched off very quickly.
How this works, is interesting. Imagine the situation where Q1 is switched ON, because the PWM
signal is high (3.3V), causing Q2 to be switched ON. D10’s cathode is pulled to ground throughQ2, and the voltage on Q2’s anode is therefore something like 0.7V. This is plenty far below the
QDX assembly Rev 1.18d

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Q1 P-channel MOSFET’s source voltage, to turn on Q1 in saturation. At the same time, the basevoltage of NPN transistor Q14 is at 0V, while its emitter is at +0.7V. The Vbe voltage is therefore 0.7V; anything under +0.7V will ensure that Q14 is OFF, conducting no current.
Now as soon as the PWM signal goes low (0V), Q2 switches off, becoming a very high resistance.
Q2 switches fast because the PWM output pin of the microcontroller is driving rail-to-rail with a low
output impedance. Now the cathode of D10 is no longer pulled to ground. The anode of D10(connected to Q1’s gate) is still at 0.7, held there initially due to the gate charge on the Q1
MOSFET, hence D10 does not conduct and is effectively out of the circuit. Current flows from the+12V input through R3, and it can flow through the base-emitter junction of Q14 now that theemitter is effectively near ground.
Now with Q14 switched on, current passes through Q14 to discharge the Q1 switch MOSFET gatecharge quickly. The current is limited only by the input current (through the 10K R3) multiplied by
the hFE of the NPN Q14 transistor. It results in a quick discharge and switch off of the Q1
MOSFET, minimizing power loss in the on/off transition.
Q14 continues to conduct while base current can flow – which is to say, while the Vbe voltage isgreater than 0.7. So it can discharge the Q1 gate capacitance right down to 0.7V from the +12V
supply; by this time the gate voltage is well below the threshold voltage at which switching occurs.
Then Q14 stops conducting and nothing more happens until the PWM signal goes high again,
switching everything ON again.
This is why it’s been called a “pseudo push-pull” drive. Really it pulls, and does not “push”; but it
does “push” momentarily, enough to efficiently rapidly switch off Q1.
Various different inductors were tried during development, and switching frequencies in the range
12kHz to 900kHz. Low switching frequency harms efficiency because a large lossy inductor isrequired; very high switching frequency harms efficiency due to the power loss at each switchtransition.

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Well there are trade-offs everywhere, but fortunately it is not particularly critical. In the final
implemented design, the inductor is 330uH and the switching occurs at approximately 100kHz.
The value of R3 also has an effect on efficiency. The lower R3 is, the more current will beavailable through Q14 to discharge the Q1 gate more rapidly on the On-to-Off transition. On theother hand, while the Q2 switch is ON, R3 is dissipating power, just wasting it; and this isminimized by using a large value for R3. The original circuit (see link above) used 2.2K but thatcircuit used a larger switching MOSFET and was designed for higher current loads. I found thatagain the value of R3 was not particularly critical but increasing it to 10K did gain a few percentage
points of efficiency.
The plot of efficiency vsinput voltage (right)
shows the normal
expected decrease inefficiency as the supply
voltage is increased andthe input-to-output voltage
ratio is therefore
increased.
Before and after close-in
spectrum analyzer
sweeps were done andno interference observed
from harmonics of the
100kHz switchingfrequency.
The harmonic attenuation was found to be several dB improved relative to the earlier circuit; I
speculate that this is because the 470uF smoothing capacitor at the buck converter output further
prevents the signal leakage path that exists past the low pass filters, through R2 and R4.
Performance
Analysis of the final Rev 4
manufactured PCB showedvery satisfactory performance.
Firmware 1_05 and above
must be used, as thisfirmware contains the control
loop for the switched mode
power supply (buckconverter). In firmware 1_05
there is an additional “PIN fwd
bias (mA) setting in the Band
Configuration screen, allowing
5-60mA and defaulting to
30mA. This is used by thecontrol loop to determine the necessary output voltage of the buck converter.
The buck converter output voltage feeds current to the two PIN diodes via resistors R2 and R4,
which are now 33-ohms. Along with the PIN diodes themselves (1N4007’s in reality), the path to
QDX assembly Rev 1.18d

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ground also contains two 47uH chokes, the toroidal inductors of the Low Pass Filter, and one of
the BS170 switch transistors which switches on the desired LPF. The microcontroller must really
strive to keep a constant current through the PIN diodes, as specified by the configuration
parameter in the Band Configuration screen. It must therefore do this by calculating the voltage
drop across the 33-ohm resistors R2 and R4. However the voltage on the downstream side of R2
and R4 is not constant. It is NOT simply the relatively constant diode voltage drop; there is also acurrent-dependent voltage drop, due to the resistive component of the load, made up of theresistance of the two 47uH chokes, toroidal inductor, and BS170 drain-source “on” resistance.
Detailed measurements
were undertaken to
separate the fixed diodevoltage drop and thecurrent-dependent
resistive drop (which
actually also neatly
includes the drop in diodevoltage drop as thecurrent changes!). A
spreadsheet was used toplot the voltage vscurrent, and draw a linear
line of best fit, determining
parameters to use in the
microcontroller code. It is also necessary to include the current contribution from the 1K resistors
R1 and R6, which feed current on both receive AND transmit from the +5V bus. This current
contribution is in the region of 4mA but again, varies depending on the total current since thevoltage drop has current dependence as per the above discussion.
There will indeed be some variation from device to device due to component tolerances but I am
satisfied that the inaccuracy will beof no importance, the performanceof the circuit is NOT marginal anddoes not depend on critical value ofcomponent choices.
Once all this was modeled in thefirmware the actual bias current vstarget bias current set in the Band
Configuration screen was carefullymeasured and they were found tomatch very closely (the red and bluelines in the chart are practicallyindistinguishable).
Next after a range of detailed
measurements on QDX units powered at 9V then at 12V, it became possible to calculate and plot
several other graphs.
First is the efficiency, which is plotted (below right). The X-axis here represents the total current
load on the SMPS, which is twice the PIN diode forward bias current since the total current isshared between two identical PIN diode circuits.

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Typically we will choose aforward bias current around
30mA (which is the default
setting in the Band Configuration
screen). This means 60mA totalload on the SMPS and theefficiency is found a little over
70%. This closely matches thevalues found during development
and also approximates theefficiencies obtained at similar
currents for several SMPS IC’sstudied.
There follows a plot of the power
dissipation vs bias current
setting, for previous PCB revision QDXs (BLUE line) and now Rev 4 (RED line).

In reality on a previous PCB the bias current was not adjustable, it was fixed by the supply voltage
and the value of resistors R2 and R4 (330-ohm). Since previous PCB revisions did not provide adecent constant current source, they merely provided forward bias current in a simple way via a
resistor from +12V (switched on during transmit); the bias current was different depending on
supply voltage. The value of R2 and R4 had to be chosen for the “worst case” and then someconsiderable safety margin, because very conceivably the QDX might be operated at lower supply
voltages even than 9V.
On a 9V QDX, the bias current was about 30mA and on a 12V QDX, about 40mA. On the Rev 4
QDX we have effectively implemented this constant current bias supply control loop and so we
can choose the lower value of 30mA. In fact, since we no longer need the safety margin (foroperating voltages below 9V) we could go even lower, until the point where spectrum analysis
shows harmonic performance deteriorates unacceptably, as diode distortion rises due to too-low
forward bias current.
For the purposes of comparison the blue line is calculated, since the PIN diode voltage drop(including switching transistor and 47uH chokes etc) is known, and displayed as the blue line,
rather than a fixed point at 40mA (12V) or 30mA (9V).
These charts show the total additional power dissipation in QDX during transmit, that is spent on
the increased forward bias diode current necessary to switch 5W through the LPF without
unacceptable distortion. They don’t include the 4mA which is present continuously, even on
receive.

QDX assembly Rev 1.18d

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Now it is possible to calculate a “before” and “after” power dissipation.
A 12V Rev1 to 3a board would run bias current per diode at about 40mA, resulting in a total
additional Transmit heat dissipation of 856mW.
The Rev 4 board, run at 30mA (constant current) has a total additional Transmit heat dissipation of
134mW.
A massive saving of 722mW, almost ¾W of heating is now removed from the QDX enclosure!
Now let’s examine the heat dissipation of each of the resistors R2 and R4:

In a 12V QDX Rev 1 to 3a, according to the measurement data, R2 dissipates 387mW.
In a 12V QDX Rev 4 at 30mA constant current forward bias current, R2 dissipates just 22mW.
So we see a massive reduction in the heat dissipation in R2 and R4, which will definitely be within
the specification of an 0603 sized resistor and will result in much lower temperature rise in thearea of these components. This is exactly the desired result from this change to the bias circuit.
Since the SMPS will only be enabled during transmit, it is necessary to examine the performanceof the SMPS as it is switched on/off on transmit/receive transitions.

These two oscillscope photographs show switch-on (left) and switch off (right). The top (yellow)
trace has a vertical scale of 500mV/div and shows the SMPS output voltage. The lower (blue)
trace is here connected to the Q1 drain at 5V/div. Since the horizontal scale is 2ms/div, very much
too slow to show the 100kHz switching waveform, the blue trace just shows here as a solid block.
It is found that at switch-on, the SMPS reaches a stable desired output voltage in approximately
2ms, and the voltage decays in about 20ms at switch-off. There are no obnoxious spikes at the
moment of switch on and switch off.

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In this trace (right) the terminal
application “Transmit Test’ is used.
The yellow trace again shows the
SMPS output at 500mV/div and thelower trace shows the Q1 drain again
at 5/div. Horizontal scale is now
100ms/div.
A single T keypress causes a 0.5
second transmission burst. When
operated from the TX Test screen, theRF burst starts almost immediately on
the key-press.
However, in a more practical scenario
using WSJT-X, the result is as shown(right). In this photograph, WSJT-X’s“Tune” button is clicked on and off
momentarily for a few seconds
approximately.
The horizontal scale is 500ms/div andthe blue channel is here connected to
the RF output across a dummy load, at
10V/div. It is evident that thetransmission from WSJT-X starts andends significantly before and after thereceive/transmit switchover, a timedelay which dwarfs the 2ms rise time of
the SMPS output voltage. In otherwords, the forward bias current is ready and waiting long before the RF arrives.
Overall it’s a very worthy modification to the QDX, providing substantial reduction in heating duringtransmit, and ensuring the R2 and R4 resistors are not operated out of specification.
To summarise:•

Constant current forward bias setting per band, configurable 5mA to 60mA (30mA default).

•

Measured reduction in transmit heating in a 12V QDX, from 856mW to 134mA (722mW
reduction, a factor of more than 6).

•

Measured reduction in R2 and R4 dissipation from 387mW to 22mW, a 17-fold reduction.

•

No impact on receive or transmit performance.

QDX assembly Rev 1.18d

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3.15 Configurable PTT (Push To Talk) output
The Rev 3 QDX PCB has a 3.5mm stereo jack as PTT output.
The “Tip” connection is pulled to ground during transmit, when Q13 is switched on by the
microcontroller.
The “Ring” connection is pulled to +5V during transmit when Q12 is switched on by the
microcontroller.
A common standard 3.5mm stereo audio cable with
3.5mm plugs at each end may be used without
modification between the QDX and the QRP Labs 50W PA(which uses the positive-going +5V signal on the “ring” andmakes no connection to “tip”.
During transmit, the microcontroller activates ONE of the
PTT signals; you may configure which one, according to
your needs.
It should be noted that:•

The QRP Labs 50W PA may not be used in highduty cycle modes such as FT8, JS8 etc at fullpower. It should be de-rated to half power operation
at maximum, by using a 12V or 13.8V supply.
Caution is advised.

•

The 50W PA kit has a built-in Low Pass Filter and isdesigned for signal band operation only.

•

Firmware 1.04 or above is required to use the PTT feature.

QDX assembly Rev 1.18d

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3.16 Voltage regulation and supply decoupling

The two fixed voltage regulators in the QDX design produce +3.3V and +5.0V. The +5V voltage
regulator type is 78M05 and the 3.3V type is AMS1117.
The 78M05 voltage regulator is robust, virtually indestructible. The AMS1117-3.3V regulator is amuch more delicate IC, but in this application it is only regulating voltage down from +5V from the
78M05, so it is not being stressed and has a very low risk of failure. Note that this is a designchange with respect to the Rev 1 QDX PCB where both voltage regulators were AMS1117,
necessitating an additional 220uF capacitor at the power input; with the 78M05 used in Rev 2 andabove, this 220uF capacitor is not needed.
PCB Rev 2 and above additionally have a reverse polarity protection feature provided by the PChannel MOSFET type AO3407/AOD403. This transistor may appear, at first sight, to beconnected “in reverse”, since the drain is connected to the incoming DC voltage from the DC
connector, and the source is connected to the downstream circuits. However, it is in fact correct
and necessary.
In normal operation, at power-up a little current will flow through the forward-biased “body diode”
of the AO3407, raising the voltage at the source pin (on the right hand side of the transistor as
drawn in the diagram). The gate is connected to half the supply voltage via R30/R35, hence thegate voltage is 6V (say) below the source voltage, and the transistor switches on, as desired.
2.2uF capacitor C55 provides smoothing for any small spikes that may be induced in the supply
wiring or from the power supply.
If the operator should happen to accidentally connect QDX in reverse, then the gate-source
voltage will be positive, which will ensure the P-channel MOSFET is switched OFF and no current
flows through it. Since the transistor was connected “backwards”, if the voltage is applied inreverse, no current flows through the “body diode” either. It therefore achieves the desired reverse
polarity protection function.
Despite the 4A continuous current rating being approximately 4x more than needed in theapplication, there were still reports of fried AO3407 MOSFETs; therefore for the Rev 4 PCB this
MOSFET was changed to AOD403 which has a 15A continuous current rating, lower ONresistance, and is packaged in a TO-252 style case that has a much higher heat dissipation
capability. This should be more robust.
QDX assembly Rev 1.18d

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4

Performance measurements

Several QDX units have been measured; there are of course variations from one to another, due
to component tolerances, toroid winding style, etc. The results presented here are for a sample
QDX; all measured QDX were reasonably similar.

4.1 Receive current consumption
The following chart shows receive current vs supply voltage, per band. There is not much variationand it is fair to quote 100mA as the receive current. For Rev 3 onward, 150mA is normal.

4.2 Transmit current consumption
The following chart shows transmit current vs supply voltage, per band.

QDX assembly Rev 1.18d

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4.3 Power output vs supply voltage
Power output is consistent across bands. Operation at more than 6W output is NOTRECOMMENDED and may destroy the power amplifier transistors.

The following graph shows power output vs supply voltage for 40m operation, with the twoalternative output transformer winding arrangements. 3:2 is less efficient but more suitable if you
need to operate from a 12 – 13 V supply.

QDX assembly Rev 1.18d

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4.4 Output harmonic content
Equipment:
QDX > QRP Labs 50-ohm dummy load kit > attenuator > Advantest R3361C Spectrum Analyzer.

Band

80m

40m

30m

20m

2nd harmonic-53 dBc-55 dBc-43 dBc-55 dBc

3rd harmonic-64 dBc-71 dBc-56 dBc-57 dBc

4th harmonic-70 dBcth

5 harmonic-70 dBc

Undetectable Undetectable-73 dBc

QDX assembly Rev 1.18d-62 dBc-70 dBc
Undetectable914.5 Unwanted sideband suppression
Measured using the QDX’s internal signal generator; unwanted sideband is 60-70 dB down.

This is also seen on the QDX’s own terminal application, Audio sweep tool:

QDX assembly Rev 1.18d

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5. Resources

For updates and tips relating to this kit please visit the QRP Labs QDX kit page http://qrplabs.com/qdx
For any questions regarding the assembly and operation of this kit please join the QRP
Labs group, see http://qrp-labs.com/group for details

6. Document Revision History
1.00

09-Oct-2021

First draft version version 1.00

1.01

27-Oct-2021

Lots of corrections; and update for firmware 1.01

1.02

27-Oct-2021

Added note about IQ Mode not being applicable to WSJT-X and digi
modes software; and that firmware update files downloaded from the
QRP Labs website need to be unzipped before installation

1.03

28-Oct-2021

Added 0.33mm wire specification to sections 2.11 and 2.12

1.04

28-Oct-2021

Clarified T1 diagram 3:3 for 9V operation; and added notes to the text
about remembering to wind 2 turns secondary for 12V operation.

1.05

08-Nov-2021

Minor typos corrected

1.06

10-Nov-2021

Added recommended comms port settings for Windows, which seem tomake CAT communications more reliable according to some user
reports.

1.07

23-Dec-2021

Revised for PCB Rev 2

1.08

29-Dec-2021

Corrected error in section 2.12, continuity checks on trifilar transformer

1.09

03-Jan-2022

Corrections to page 24 diagram, removed R34 reference, and typo on
page 17 (thanks Dave M0JTS for pointing these out)

1.10

06-Jan-2022

Re-format only to correct some formatting problems

1.11

07-May-2022

Revised for PCB Rev 3

1.11a 27-Jul-2022

Corrected typo, Q7 in parts list is AO3407 (4307 was wrong)

1.12

03-Aug-2022

Revised for PCB Rev 3a (section 2.3a particularly, and no need for the
insulating tape at section 2.15)

1.13

15-Sep-2022

Corrected typo, inductors are axial not radial (thanks David KC3AWJ)

1.14

30-Sep-2022

Revised for PCB Rev 4

1.15

19-Oct-2022

1.16

23-Dec-2022

Revised for high-bands QDX version

1.17

30-Dec-2022

Revised for PCB Rev 5

1.18

09-Jan-2023

Fix typo on page 31

1.18a 07-Feb-2023

Fix 0.3mm wire length for 20-10m version in parts list
QDX assembly Rev 1.18d

93

1.18b 15-Feb-2023

Added comment about D1/L3 connection in section 2.6

1.18c 20-Feb-2023

Section 2.9 typo on number of turns for longest part of winding for highbands version

1.18d 24-Feb-2023

Corrected footer

QDX assembly Rev 1.18d

94