550

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 3, MARCH 2010

Ripple Current Reduction of a Fuel Cellfor a Single-Phase Isolated Converter Usinga DC Active Filter With a Center Tap
Jun-ichi Itoh, Member, IEEE, and Fumihiro Hayashi

Abstract—A ripple current reduction method is proposed that
does not require additional switching devices. A current ripple thathas twice the frequency component of the power supply is generated in the dc part when a single-phase pulsewidth-modulated
inverter is used for a grid connection. The current ripple causes
shortening of the lifetime of electrolytic capacitors, batteries, andfuel cells. The proposed circuit realizes a dc active filter function
without increasing the number of switching devices, because the energy buffer capacitor is connected to the center tap of the isolationtransformer. In addition, the buffer capacitor voltage is controlled
by the common-mode voltage of the inverter. The features of the
proposed circuit, control strategy, and experimental results are described, including the result of ripple reduction, to approximately
20% that of the conventional circuit.
Index Terms—Center tap, dc active filter, grid connection system, ripple current reduction, single-phase isolated converter.

I. INTRODUCTIONECENTLY, energy sources such as wind power systems,
photovoltaic cells, and fuel cells have been extensively
studied in response to global warming and environmental issues. The fuel cell is an important technology for new mobile
applications and power grid distribution systems. For power distribution, fuel cell system requires a grid interconnection converter to supply power to the power grid. A grid interconnection
converter using an isolation transformer is preferable for power
grid distribution systems in terms of surge protection and noise
reduction. In addition, size reduction and high efficiency are
essential requirements [1]–[8].
One of the problems in the fuel cell system is that the lifetime
is decreased by the ripple current. Therefore, in order to extend
the lifetime, the fuel cell ripple current must be reduced in
the grid interconnection converter [9], [10]. However, when asingle-phase pulsewidth-modulated (PWM) inverter is used forgrid connection system, the power ripple is twice the frequencyof the power grid. Therefore, in conventional grid connection
inverters, large electrolytic capacitors are connected in parallelto the fuel cell in order to reduce the current ripple. However,R

Manuscript received March 30, 2009; revised June 10, 2009. Current version
published March 31, 2010. A part of this paper was presented at the IEEE Applied Power Electronics Conference 2009, Washington, DC. Recommended for
publication by Associate Editor S. Choi.
J.-i. Itoh is with the Nagaoka University of Technology, Nagaoka, Niigata
940-2188, Japan (e-mail: itoh@vos.nagaokaut.ac.jp).
F. Hayashi was with the Nagaoka University of Technology, Nagaoka,
Niigata 940-2188, Japan. He is now with Shikoku Electric Power Company,
Inc., Kagawa 761-8573, Japan (e-mail: bunpaku@stn.nagaokaut.ac.jp).
Digital Object Identifier 10.1109/TPEL.2009.2029106

the use of large-sized electrolytic capacitors increases both thedevice volume and cost.
In order to reduce the current ripple in the fuel cell, some approaches use high-speed current control [11], [12]. This method
incorporates a current-loop control within the existing dc–dc
converter voltage loop. However, a large capacitor or reactor is required as an energy buffer. Other approaches havebeen proposed that do not require the use of large-sized electrolytic capacitors, e.g., an active filter is applied in the dc-linkpart [13]–[16]. The dc active filter consists of a small capacitor
as an energy buffer, a reactor to reduce the switching ripple, anda dc chopper. The dc chopper injects the ripple current to avoid apower ripple. The capacitance can be lower, because the terminal
voltage of the capacitor can be varied over a wide range. However, the number of the switching devices is increased, requiringa high-cost dc chopper and resulting in a large volume device.
Other configurations of dc active filters have similar problems.
This paper proposes a new circuit topology including a dc
active filter function without extra switching devices. The proposed circuit consists of an isolated dc/dc converter and interconnection inverter, and achieves the dc active filter function using the center tap of the isolation transformer. One feature of the
proposed converter is that the primary-side inverter in the dc/dc
converter is individually controlled by the common-mode voltage and the differential voltage. The ripple current is suppressedby the common-mode voltage control of the dc/dc converter,and the main power flow is controlled by the differential-mode
voltage.
Conventional and proposed circuit topologies with the principle of current ripple suppression are first introduced. The control
method of the proposed circuit is then described. In addition, thedesign of the energy buffer capacitor and transformer by which
the maximum power ripple can be accepted is indicated. Furthermore, experimental results are presented in order to confirm
the validity of the proposed circuit.
II. PROPOSED CIRCUIT CONFIGURATIONS
Fig. 1 shows a conventional circuit that consists of a firststage inverter for the medium frequency link, a transformer, adiode rectifier, and a grid interconnection inverter. When theinterconnection current and power grid voltage are sinusoidal
waveforms, the instantaneous power p of the grid interconnection is obtained by (1) at unity power factor√√
p = 2I sin(ωt) 2V sin(ωt)= VI{1 − cos(2ωt)}(1)0885-8993/$26.00 © 2010 IEEE
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ITOH AND HAYASHI: RIPPLE CURRENT REDUCTION OF A FUEL CELL FOR A SINGLE-PHASE ISOLATED CONVERTER

551

TABLE ICOMPARISON OF THE NUMBER OF SWITCHING DEVICES
AND THE CAPACITOR CAPACITY

Fig. 1.

Fig. 2.

Conventional circuit.

Conventional circuit with dc active filter.

capacitor Cf is connected to the center tap of a mediumfrequency transformer. The zero vector of the full-bridge firststage inverter is used to control the center tap potential voltage.
In addition, the leakage inductance of the transformer is used tosuppress the switching current in addition to the boost reactor
Lf . If the leakage inductance is large enough, then the boost
reactor is not required.
Table I provides a comparison of the number of switching
devices and the capacitor capacity of a conventional circuit, aconventional circuit with a dc active filter, and the proposed circuit. The proposed circuit does not require additional switching
devices or an inductor, in comparison to the conventional circuitwith the dc active filter. It should be noted that the current ratingsof the switching devices are larger than that of the conventional
circuit, depending on the capacity of the energy buffer, because
the dc active filter current flows in the first-stage inverter andthe transformer.
III. CONTROL STRATEGY

Fig. 3.

Proposed circuit.

where I and V are the rms values of the interconnection current
and the grid voltage, and ω is the grid angular frequency.
Thus, the instantaneous power has a ripple that is twice thefrequency of the power grid frequency. To reduce the ripple
power of a dc power source, such as a fuel cell, battery, or
photovoltaic cell, large electrolytic capacitors Cdc1 and Cdc2
are used in the converter, as shown in Fig. 1. The use of large
electrolytic capacitors precludes reduction in size and cost.
Fig. 2 shows the other conventional circuit using a dc active
filter, which is constructed using a dc chopper and an energy
buffer capacitor Cf . The capacitor Cf is used as an energy buffer
to absorb the ripple power. The inductor Lf can suppress theswitching current. The voltage of the capacitor Cf is controlled
at twice the frequency of the power grid frequency. As a result,
the ripple power does not appear in Vdc , despite the use of small
capacitors Cdc1 and Cdc2 . However, the problem of this method
is that the number of switching elements is increased.
Fig. 3 shows the proposed circuit, which combines the firststage inverter and dc active filter functions. The energy buffer

The first-stage inverter in the proposed circuit has two roles
to perform: that of a dc/dc converter and that of a dc active filter.
These roles are achieved by controlling the common-mode anddifferential-mode voltages in the first-stage inverter. This paper
explains the principle of the proposed control method and thedesign method for the buffer capacitor and the transformer.
A. Switching Pattern Generation Method
Fig. 4 illustrates the two switching modes of the first-stage
inverter in the proposed circuit. In the differential mode, theterminal voltage of the transformer is controlled, as shown in
Fig. 4(a) and (b), and in the common mode, the center tap voltage
is controlled, as shown in Fig. 4(c) and (d). When the differential
mode is selected, the power transfers to the secondary side. It
should be noted that the buffer current for the differential mode
depends on the capacitor voltage. When the capacitor voltage
is greater than half of the dc voltage, the buffer capacitor ischarged. When the capacitor voltage is less than half of the dc
voltage, the buffer capacitor is discharged. The inverter outputs
the zero voltage vectors (00 and 11 are two) in common-mode
operation. When the zero voltage vectors are selected, the lineto-line voltage of the transformer is zero. However, the centertap voltage is either Vdc or zero, depending on the zero vectors of
Fig. 4(c) or (d), respectively. Thus, by controlling the ratio of thezero vectors, the buffer capacitor can be charged or discharged.

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552

Fig. 4.

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 3, MARCH 2010

Operation modes of the proposed circuit. (a) Differential mode 1. (b) Differential mode 2. (c) Common mode 1. (d) Common mode 2.

obtained by


1 ∗

∗
 v1∗ = (vcom+ vdif
)

2


∗
∗
 v2∗ = 1 (vcom− vdif).
2(2)

Grid interconnection control can be applied to the conventional control method, which uses an automatic current regulator∗. The in(ACR) for the interconnection current command Igrid
terconnection current command is obtained by the synchronous
signal to the grid voltage.
Fig. 5. Current and voltage waveforms for differential-mode and commonmode operations.

It should be noted that the output-switching pattern must include
the zero vector period. Therefore, the voltage transfer ratio of
the first-stage inverter is limited by the dc active filter control.
As a result, the terminal voltage of the transformer is decreased.
Fig. 5 illustrates the operation waveforms with consideration of the transformer. The waveforms confirm that the transformer current consists of the common-mode current, whichis the buffer current, and the differential-mode current. The
common-mode current increases while S1 and S2 are turned on,
and decreases during differential mode, because the capacitor
voltage is greater than half of the dc voltage.
Fig. 6 shows a control block diagram of the proposed circuit. In order to suppress the ripple current of the fuel cell, all
the current ripples are provided by the energy buffer capacitor.
Therefore, the capacitor current command i∗com is obtained by
calculating the power ripple. The capacitor current icom in theactive buffer is controlled by a proportional–integral (PI) regulator to agree with the capacitor current command, and the dc∗
is output by the PI regulator.
active filter voltage command vcom
In order to obtain the maximum terminal voltage of the trans∗
is set to theformer, the differential-mode voltage command vdif
maximum value. It should be noted that the maximum value ofthe voltage command is “1” when the peak value of the triangle
carrier is “1.”The feature of the proposed circuit control is that the dc active∗is added to the differential voltage
filter voltage command vcom∗
command vdif as the common-mode voltage. The output voltage
commands v1∗ and v2∗ for each leg in the first-stage inverter are

B. Design of the Buffer Capacitor for dc Active Filter Operation
The buffer capacitor is used as an energy storage element of
the active filter. The capacitor Cf has to absorb the power ripple
for a half cycle of the power grid. Thus, the required storage
energy WC is given by (3) from (1) and the capacitor energy
Wf is obtained by (4)
 T/4
Pin
(3)
Pin cos 2ωtdt =
WC =
ω
0


1
Wf = Cf Vf2m ax − Vf2m in(4)
2
where Pin is the input power, and Vf m ax and Vf m in are themaximum and minimum voltages of Cf , respectively.
Therefore, the required capacitance Cf is given by (5) from
(3) and (4) as
Cf =
=

2Pin
ω((Vc f 0 + ∆Vc f /2)2 − (Vc f 0 − ∆Vc f /2)2 )Pin
ωVc f 0 ∆Vc f(5)

where Vc f 0 and ∆Vc f are the average voltage and the voltage
variation of Cf , respectively.
Fig. 7 shows the relation between the capacitance and ∆Vc f
of the capacitor to compensate the power ripple at 1 kW and
Vc f 0 of 150 V, according to (5). The capacitance can be significantly reduced by the capacitor voltage variation. It should be
noted that the percentage impedance of the 1000 µF capacitor
reciprocates with 3.18 p.u. based on 200 V, 50 Hz, and 1 kW.
The reactor Lf is set to decrease the switching ripple, i.e., Lf
depends on the switching frequency of the first-stage inverter;
the leakage inductance of the transformer is used as Lf .

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ITOH AND HAYASHI: RIPPLE CURRENT REDUCTION OF A FUEL CELL FOR A SINGLE-PHASE ISOLATED CONVERTER

Fig. 6.

553

Control block diagram of the proposed circuit.

Fig. 8. Current and voltage wave pattern according to the difference in thecommon mode duty ratio at constant power.
Fig. 7. Required capacitance for power ripple compensation. (P in = 1 kW,
V d c = 300 V, V c f 0 = 150 V, 1000 µF = 3.18 p.u.).

C. Design of the Transformer
The major focus in the design for the transformer is the current
capacity. This is because the transformer has two functions: dc
active filter and isolation transformer. The transformer current
Itran is equaled to the sum of the active filter current Icom andthe current Idif , according to the output power shown in
Icom
.
(6)
2
Note that in (6), the active filter current is divided by 2,
because the transformer winding is connected in parallel to thecenter tap in the common-mode circuit, as shown in Fig. 4(c)and (d).
The common-mode voltage controls the capacitor voltage
variation, and the differential-mode voltage controls the transmission power to the power grid. The common-mode voltage
should be varied widely to suppress the capacitance, as shownin (5). However, the output period of the common-mode voltage
is limited by the output period of the differential mode, i.e., theduty ratio Ddif for the differential mode can be constrained by
Itrans = Idif +Ddif + Dcom = 1

output power. The differential-mode current increases when thecommon-mode voltage increases, because the power transmission period is reduced by the common-mode voltage. Therefore,a large common mode voltage causes a large current in the transformer. The transformer current is then obtained as the next step.
The duty ratio for the common-mode voltage is obtained by(8), because the average voltage Vc f 0 of the capacitor Cf is halfof the dc voltage used to achieve the maximum capacitor voltage
variation
∆Vc f
∆Vc f
Dcom ==(8)
2Vc f 0
Vdc
where Vdc is the dc voltage of the first-stage inverter.
The differential-mode current Idif is then obtained using (9)from (7) and (8) as
Idif =Pin
Pin=.
Vdc Ddif m ax
Vdc − ∆Vc f

On the other hand, the common-mode current Icom is definedby (10), because the maximum value of the power ripple is twicethat of the input power
Icom =(7)

where Dcom is the duty ratio for the common-mode voltage.
Fig. 8 shows the differential-mode current waveforms andthe terminal voltage waveforms of the transformer when thecommon-mode command is 0 and 0.3 p.u., given at the same

(9)

2Pin
.
Vdc

(10)

Finally, the transformer current Itrans is obtained by (11) from
(6), (9), and (10) as
Itrans = Idif +

Pin
Icom
Pin
=
+
.
2
Vdc − ∆Vc f
Vdc

(11)

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554

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TABLE II
EXPERIMENTAL PARAMETERS WITH PERCENTAGE IMPEDANCE SHOWNIN PARENTHESES

Fig. 9. Current capacity of the transformer necessary for compensation (P in =1 kW, V d c = 300 V, V c f 0 = 150 V).

Fig. 10. Current capacity and capacitance of the transformer necessary
for compensation (P in = 1 kW, V d c = 300 V, V c f 0 = 150 V, 1000 µF =3.18 p.u).

method requires a fat winding transformer compared with that
required for the conventional circuit.
Note that the number of turns in the transformer winding is
calculated from (14), as for the conventional transformerN=

It should be noted that Icom in (10) is the peak value of thecommon-mode current. Therefore, the rms value Itrans(rm s) of
the transformer current, which is used to determine the thicknessof the transformer winding, is obtained by
Itrans(rm s) = Idif +

Pin
Icom
Pin
=
+√
.
2
Vdc − ∆Vc f
2Vdc

(12)

Fig. 9 presents the relation between the transformer current and the capacitor voltage variation at Vc f 0 = 150 V, Pin =1 kW, and Vdc = 300 V. As the capacitor voltage variation increases, a smaller active filter capacitor Cf can be achieved.
However, as the capacitor voltage variation becomes larger, it
results in a system that requires a large current capacity transformer. It should be noted that the current rating of the switching
device is also dominated by (11).
In addition, the relation between the transformer current andthe capacitance is represented by (13), from (5) and (11), asCf =Pin (Pin − Vdc Itrans ).
ωVc f 0 Vdc (2Pin − Vdc Itrans )(13)

Fig. 10 further illustrates the relation between the transformer
current and the capacitance for Vc f 0 = 150 V, Pin = 1 kW, and
Vdc = 300 V. Fig. 10 shows that selection of a smaller capacitance results in the requirement of a larger current capacitytransformer. For example, if a 150-µF capacitor is selected fora 1-kW system, then a transformer with a current capacity ofmore than 10 A must be considered. In other words, the proposed

Vdc
4f SB

(14)

where N is the number of turns in the winding, f is the switchingfrequency of the first-stage inverter, B is the flux density of thecore, and S is the section area of the core.
IV. EXPERIMENTAL RESULTS
The proposed converter was tested to confirm the validityof the proposed circuit operation using the experimental conditions given in Table II. The proposed control shown in Fig. 6 is
implemented using a digital control system with a digital signal processor (DSP). The auxiliary inductor is connected to thecenter tap of the transformer, because the leakage inductance ofthe transformer is not sufficient to reduce the switching ripple
current.
Fig. 11 shows the operation waveforms of the conventional
circuit without a dc active filter. A sinusoidal grid current waveform and unity power factor are obtained; however, the dc input
current has a large ripple current component of 100 Hz.
Fig. 12 shows the operation waveforms of the proposed converter. The ripple of the dc input current is suppressed to 20%
that of the conventional circuit, which indicates that the dc active
filter function is effective. The grid current waveform maintains
the sinusoidal waveform and unity power factor.
Fig. 13 shows the operation waveforms of the conventional
circuit using a large electrolytic capacitor of 2200 µF. This
circuit was tested in order to determine the reduction in the dc
ripple current using a large electrolytic capacitor. Although the

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ITOH AND HAYASHI: RIPPLE CURRENT REDUCTION OF A FUEL CELL FOR A SINGLE-PHASE ISOLATED CONVERTER

Fig. 11.
filter.

Operation waveforms of the conventional circuit without a dc active

Fig. 12.

Operation waveforms of the proposed circuit.

Fig. 13. Operation waveforms of the conventional circuit with a large electrolytic capacitor C d c 2 = 2200 µF.

large electrolytic capacitor used is 20 times than that shown in
Fig. 12, the dc input current was not significantly reduced.
Fig. 14 shows the dc input current total harmonic distortion(THD) of the conventional and proposed circuit, which is defined by (14)

In2
(15)
Idc THD =Idc
where In is the harmonic components and Idc is the dc
component.
The major harmonic component in the input current is 100 Hz.
In a conventional circuit, the dc input current THD increases
according to the increment of the output power. In contrast, thedc input current THD decreases despite the increment of theoutput power in the proposed circuit, i.e., the proposed circuit

Fig. 14.

555

Input dc current distortion factor.

Fig. 15. THD of the grid connection current for the conventional and proposed
circuits.

Fig. 16.

Efficiency and grid connection power factor.

is suitable for high-power applications, due to its effectivenessin the high-output power region.
Fig. 15 shows the THD of the grid interconnection current forthe conventional and proposed circuits; almost the same current
THD values were obtained. Therefore, the proposed circuit canachieve the same performance level as the conventional circuit.
Fig. 16 shows the efficiency and the grid interconnection
power factor of the conventional and proposed circuits. The
efficiency of the conventional circuit is higher than the proposed
circuit. One of the reasons for the increase in power loss is theincreasing current in the transformer. Therefore, the efficiencyof the proposed circuit can be improved if the design of thetransformer is optimized. Note that the proposed converter has
good performance as a grid interconnection converter, because
both power factors of the proposed circuit and the conventional
circuit are 99%.

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556

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The experimental results confirm that the proposed converter
is valid for the reduction of the dc input ripple current in a dc
power supply, without the need for large electrolytic capacitors.

[9] D. Polenov, H. Mehlich, and J. Lutz, “Requirements for MOSFETs in fuelcell power conditioning applications,” in Proc. IEEE EPE-PEMC 2006,
pp. 1974–1979.
[10] G. Fontes, C. Turpin, S. Astier, and T. A. Meynard, “Interactions between
fuel cells and power converters: Influence of current harmonics on a fuelcell stack,” IEEE Trans. Power Electron., vol. 22, no. 2, pp. 670–678,
Mar. 2007.
[11] C. Liu and J.-S. Lai, “Low frequency current ripple reduction techniquewith active control in a fuel cell power system with inverter load,” IEEE
Trans. Power Electron., vol. 22, no. 4, pp. 1429–1436, Jul. 2007.
[12] S. K. Mazumder, R. K. Burra, and K. Acharya, “A ripple-mitigating andenergy-efficient fuel cell power-conditioning system,” IEEE Trans. PowerElectron., vol. 22, no. 4, pp. 1437–1452, Jul. 2007.
[13] F. Perfumo, A. Tenconi, M. Cerchio, R. Bojoi, and G. Gianolio, “Fuelcell for electric power generation: Peculiarities and dedicated solutions
for power electronic conditioning systems,” EPE J., vol. 16, pp. 44–50,
Feb. 2006.
[14] M. Pereira, G. Wild, H. Huang, and K. Sadek, “Active filters in HVDC
systems: Actual concepts and application experience,” in Proc. Int. Conf.
Power Syst. Technol. 2002. (PowerCon), pp. 989–993.
[15] M. Saito and N. Matsui, “Modeling and control strategy for a single-phase
PWM rectifier using a single-phase instantaneous active/reactive power
theory,” in Proc. IEEE INTELEC 2003, pp. 573–578.
[16] X. Ma, B. Wang, F. Zhao, G. Qu, D. Gao, and Z Zhou, “A high power
low ripple high dynamic performance DC power supply based on thyristor
converter and active filter,” in Proc. IEEE IECON 2002, pp. 1238–1242.

V. CONCLUSION
A novel single-phase isolated converter was proposed for gridinterconnection applications. The ripple current in a dc power
supply, such as a fuel cell, battery, or photovoltaic cell, can be
reduced by the appropriate operation of a dc active filter. The
main feature of the proposed circuit is that it does not require
additional switching devices, because the zero vector of thefirst-stage inverter is controlled as the dc active filter. A 1-kW
prototype was constructed based on the proposed circuit, andthe following experimental results were obtained.
1) The ripple current can be decreased to 20% that of theconventional circuit.
2) The proposed circuit shows a degree of effectiveness forhigh output power applications.
3) The total electrolytic capacitor requirement of the system
is decreased to 25% that of the conventional circuit.
4) The dc active filter operation in the proposed method circuit does not interfere with the grid interconnection current
control.
In future work, optimization of the transformer and construction of a high-power prototype will be undertaken.
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Jun-ichi Itoh (A’04–M’04) was born in Tokyo,
Japan, in 1972. He received the M.S. and Ph.D. degrees in electrical and electronic systems engineeringfrom Nagaoka University of Technology, Nagaoka,
Japan, in 1996 and 2000, respectively.
Form 1996 to 2004, he was with Fuji Electric
Corporate Research and Development, Ltd., Tokyo,
Japan. Since 2004, he has been an Associate Professor in Nagaoka University of Technology, Niigata,Japan. His current research interests include matrix
converters, dc/dc converters, power factor correction
techniques, and motor drives.
Dr. Itoh is a member of the Institute of Electrical Engineers of Japan. He
received the Institute of Electrical Engineers of Japan (IEEJ) Academic Promotion Award (IEEJ Technical Development Award) in 2007.

Fumihiro Hayashi was born in Kagawa, Japan, in
1985. He received the B.S. and M.S. degrees in electrical, electronics and information engineering from
Nagaoka University of Technology, Niigata, Japan,
in 2007 and 2009, respectively.
Since 2009, he has been with Shikoku ElectricPower Company, Inc., Kagawa, Japan. His current
research interests include power grid distribution systems and new converter topologies.

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