D ata
Acquisition
Handbook

A Reference For DAQ And
Analog & Digital
Signal Conditioning

A/D

Conversion•

Accelerometers

Cold-Junction

Compensation

Displacement&

Frequency

Measurement

Instrumentation

Position

Amplifiers•

D/A

Sensing••••

Amplification

Attenuation

Conversion•

Electrical

Measurements

Fundamental

Isolation•

Digital•SignalMultiplexing•

I/O

Interfacing•

Filtering

Conditioning
Noise

Reduction

RMS Measurements • RTDs • Sensors • Strain Gages • Thermistors
T h e r m o c o u p l e s • Tr a n s d u c e r E l e c t r o n i c D a t a S h e e t s • Tr a n s d u c e r s • V i b r a t i o n

D ataAcquisition
Handbook

A Reference For DAQ And
Analog & Digital
Signal ConditioningT h i r d

E d i t i o niiCopyright Notice© 2004-2012 by Measurement Computing Corporation
All rights reserved. No part of this book may be reproduced in any form,or by any means, without the permission of Measurement Computing Corporation.
Published 2004-2012 in the United States of America

iii

Table of Contents

TABLE OF CONTENTS. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .  iiiPREFACE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .  ix
Chapter 1

Chapter 2

Introduction to Data Acquisition and Signal Conditioning . . . . . . 1
		Signals, Sensors, and Signal Conditioning . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

		Data Acquisition Systems  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

		PC-Based Data Acquisition Equipment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Analog to Digital Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
ADC Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Successive-Approximation ADCs .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 3Voltage-to-Frequency ADCs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Integrating ADCs: Dual Slope .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 4Sigma-Delta ADCs .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 5Accuracy and Resolution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Accuracy vs. Resolution . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 6ADC Accuracy vs. System Accuracy .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 8ADC Noise Histograms  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
ENOB: Effective Number of Bits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
The ENOB Test .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 10
ADC Output Averaging Benefits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Improved Accuracy .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 11
More Stable Readings . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 11

Chapter 3

Multiplexing AND SAMPLING THEORY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
The Economy of Multiplexing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Sampled-Data Systems . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 13
Solid-State Switches vs. Relays .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 13
Speed . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 14
Sequence vs. Software-Selectable Ranges .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 14

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iv

TABLE OF CONTENTS

Fundamental Concepts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Sampling Rates  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 15
Source Impedance  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 16
Sample-and-Hold ADCs  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 16
Nyquist Theorem .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 17
Aliasing and Fourier Transforms .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 17
Settling Time .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 21
Chapter 4

Electrical Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Safety Concerns . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Making Valid Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

		DC Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Instrumentation Level DC Voltage . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 23
High DC Voltage  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 23
Low DC Voltage .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 24
AC Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Low AC Voltage .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 24
Outlet Level AC Voltage  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 24
High Voltage  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 24
Effective, Peak, Average, and RMS Voltage .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 25
Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
A Fundamental Problem .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 26
Insertion Loss .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 27
Common-Mode Limitations  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 27
Shunts .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 27
Current Transformers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 28
Hall-Effect Sensors  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 29
Resistance  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Basic Measurements . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 30
Voltage Divider Method .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 31
Constant Current/Voltage Drop  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 31
Small Deviations  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 31
Wheatstone Bridges  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Basic Principle  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 32
Anderson Loop Alternative .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 32
Single-Ended and Differential Measurements . . . . . . . . . . . . . . . . . . . . . . . 34
Basic Principle  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 34
When to Make Single-Ended Measurements .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 34
When to Make Differential Measurements  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 34
High Common-Mode Voltage Measurements  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 36
Kelvin Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

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Basic Principle  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 37
Examples of Kelvin-Type Measurements  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 37
Chapter 5

Fundamental Signal Conditioning  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Amplification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Data Acquisition Front Ends .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 39
Operational Amplifiers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 40
Inverting Amplifier Stages . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 41
Non-Inverting Amplifier Stages .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 42
Differential Amplifiers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 42
Programmable-Gain Amplifiers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 43
Instrumentation Amplifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
A Fundamental Problem .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 44
High Common-Mode Amplifiers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 44
Integrated Instrumentation Amplifiers . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 47
Programmable-Gain Instrumentation Amplifiers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 47
Filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
Low-Pass Filters  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 48
High-Pass Filters .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 49
Passive vs. Active Filters  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 49
Switched-Capacitance Filters . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 50
Attenuation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
Voltage Dividers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 51
Buffered Voltage Dividers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 52
Balanced Differential Dividers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 52
High-Voltage Dividers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 52
Compensated Voltage Dividers and Probes .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 52
Isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
When Isolation is Required .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 53
Isolation Amplifiers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 53
Analog Isolation Modules .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 54
Digital Method of Isolation . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 56
Inherently Isolated Sensors .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 57
Linearization  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60
Why Linearization is Needed  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 60
Software Linearization .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 61
Hardware Linearization  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 62
Circuit Protection  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
Hazards to Instrumentation Circuits .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 63
Overload Protection .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 64
ESD Protection .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 64

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TABLE OF CONTENTS

Chapter 6

Temperature Measurement  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
Thermocouple Basics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
The Gradient Nature of Thermocouples  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 65
Cold-Junction Compensation  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 66
Type Mixing .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 68
Thermocouple Measurement Pitfalls  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70
Noisy Environments .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 70
Additional Concerns .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 71
RTD Measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
Basics of Resistance Temperature Detectors . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 73
Measurement Approaches – 2, 3, and 4-Wire Configurations .  .  .  .  .  .  .  .  .  .  .  .  . 74
Thermistor Measurements  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78
Basics of Thermistors  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 78
Linearization  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 78
Measurement Approaches – Temperature Measurement .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 80

Chapter 7

Strain Measurements  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Strain Basics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Strain Measurement Configurations  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Wheatstone Bridge .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 81
Full-Bridge Circuits .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 82
Half-Bridge Circuits . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 83
Quarter-Bridge Circuits .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 83
Calibration Approaches  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87
Transducers and Load Cells  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 87
Strain Diaphragm Pressure Gages  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 87

Chapter 8	Vibration and Sound . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89
Sensors for Vibration and Sound Measurements . . . . . . . . . . . . . . . . . . . . . 89
Strain-Gage Accelerometers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 89
Piezoelectric Transducers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 90
Low-Impedance Transducers . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 91
Chapter 9	Displacement and Position Sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95
Linear Displacement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95
String-Potentiometer Transducers .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 95
LVDT: Linear Variable Differential Transformers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 96
Magnetostrictive Position Sensor  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 97
Linear Encoders  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 98
Rotational Position  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98
Absolute Encoders  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 99

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Incremental Encoders .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 99
Quadrature Encoders .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 100
Errors .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 100
Chapter 10

Noise Reduction and Isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
Controlling Noise  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
The Grounding Conflict  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
Grounding for Safety  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 103
Grounding for Robust Instrumentation .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 104
Ground Loops . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104
Symptoms of Ground Loops  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105
How Ground Loops Are Created .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 105
How to Eliminate Ground Loops .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 105
Crosstalk in Data Acquisition Systems  . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106
Static Crosstalk  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 106
Dynamic Crosstalk .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 106
Shielded Wiring . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106
Benefits  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 106
Proper Installation and Use of Shields . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 107
Isolation and Floating Data Acquisition Systems . . . . . . . . . . . . . . . . . . . . 108
Isolation . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 108
Circuit Protection .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 108
Rejection of High Common-Mode Voltage . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 108
Fundamental Application Mistakes .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 110
Isolation Transformers  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 111
Analog Isolators .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 112
Wireless Techniques  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 112
Noise Reduction  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113
Signal Averaging  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 113
Analog Filtering  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 113
Passive vs. Active Filters  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 114
Differential Voltage Measurements .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 114

Chapter 11	Digital and Pulse-Train Conditioning . . . . . . . . . . . . . . . . . . . . . . . . . . 117
Digital I/O Interfacing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117
Digital Signals  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 117
High Current and Voltage Digital I/O .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 117
Digital Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118
Asynchronous Digital-Input Readings .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 118
Synchronous Digital-Input Readings .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 119
Externally Triggered Digital-Input Readings . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 119

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TABLE OF CONTENTS

Digital Isolation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
Pulse Train Signal Conditioning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119
Frequency-to-Voltage Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120
Pulse-Train Integration .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 120
Digital-Pulse Counting .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 121
Frequency Measurement by Gated-Pulse Counting .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 123
Timing Applications  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 123
Chapter 12

Transducer Electronic Data Sheets . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125
Smart Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125
Networks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126
System Architecture  . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127
Part 1: NCAP Information Model  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 127
Part 2: Transducer to Microprocessor Protocol . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 127
Part 3: Digital Communications and Format . .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 127
Part 4: Mixed-Mode Communications Protocol .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 127
Part 5: Wireless Communications Protocol  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  .  . 127
TEDS Composition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127

Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129

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ix

Preface to the Third Edition

The third edition of this handbook has been
totally revised to include new chapters on
Electrical Measurements, Vibration and Sound,
Displacement and Position Sensing, andTransducer Electronic Data Sheets (TEDS). It
also includes several new subjects and expands
on selected items including Fundamental Signal
Conditioning.
All chapters have been enhanced to address
more practical applications than theoreticalmeasurement issues. They cover a major topic
with sufficient detail to help readers understand the basic principles of sensor operationand the need for careful system interconnections. The handbook also discusses key issues
concerning the data acquisition system’s
multiplexing and signal conditioning circuits,
and analog-to-digital converters. These threefunctions establish the overall accuracy, resolution, speed, and sensitivity of data acquisition systems and determine how well thesystems perform.
Data acquisition systems measure, store,
display, and analyze information collectedfrom a variety of devices. Most measurements
require a transducer or a sensor, a device thatconverts a measurable physical quantity into
an electrical signal. Examples include temperature, strain, acceleration, pressure, vibration,and sound. Yet others are humidity, flow,level, velocity, charge, pH, and chemicalcomposition.

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Sensors come in numerous shapes, sizes, andspecifications. They connect between themeasured physical device and the signal conditioner’s input. Most sensors are purchased offthe-shelf, but in some cases, they are custom
made specifically for a particular measurement
requirement. Regardless of input, however, theoutput signal is usually a voltage, current,
charge, or resistance and all can be conditioned
and handled equally well. Manufacturers
frequently provide specifications, application
notes, and principles of operation for their
specific sensor to help users apply the device inthe most efficient way.
Signal conditioners accept sensor output signals
and convert them into a form that the dataacquisition system can manipulate. Signal
conditioners typically amplify, filter, isolate,and linearize these signals. They also convert
current to voltage and voltage to frequency,
provide other functions such as simultaneous
sample and hold (SS&H), and supply a bias
voltage or signal excitation for certain transducers. They may come with single-ended
inputs or differential inputs for improving
signal-to-noise ratios. The output of the signal
conditioner, in turn, connects to the input of ananalog-to-digital converter (ADC) embedded
within the data acquisition system. Finally, theADC converts the conditioned analog signal toa digital signal that can be transferred out of thedata acquisition system to a computer forprocessing, graphing, and storing.

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x

PREFACE

Introduction to Data Acquisition and
Signal Conditioning

Chapter 1 discusses signals, sensors, andsignal-conditioning techniques and how they
relate to data acquisition system fundamentals. It also covers personal computers andhow laptop or notebook computers work withdata acquisition systems.

Analog-to-Digital Conversion.

Chapter 2 discusses four basic ADC types,
including their accuracy and resolution.
Also covered are topics such as ADC output
averaging, discrete sampling, input andsource impedance, and differential voltage
measurements. Yet others include simultaneous sample and hold methods, selectable
input ranges, aliasing, digital filtering, andFourier Transforms.

Multiplexing and Sampling Theory

Chapter 3 covers the fundamental principles ofmultiplexing and their benefits and economies.

Electrical Measurements

Chapter 4 and Chapters 6 through 9 discussbasic electrical measurements, the characteristics of various sensors, how to use sensors
to measure the most common types of electrical and physical quantities, and the signalsthat they represent. Topics include voltage,current, resistance, charge, temperature,strain, position, velocity, acceleration, andsound. The sensors that produce theminclude thermocouples, RTDs, thermistors,
strain gages, accelerometers, and linear androtational displacement sensors.

Measurement Computing(508) 946-5100

Fundamental Signal Conditioning

Chapters 5 and 11 discuss the most widelyused techniques for analog, digital, and pulsetrain signal conditioning. They comprise operational, differential, and high-gain amplifiers
for filtering, attenuation, isolation, linearization, and circuit protection for analog signals.
They also cover topics on digital I/O interfacing, frequency measurements, and patterngeneration for digital signals.

Noise Reduction and Isolation

Chapter 10 is dedicated to simplifying thesomewhat difficult topic of electrical noiseinterference, using the best shielding andgrounding techniques, and identifying the
major sources of crosstalk. It also discusses how
to select the proper amplifiers and sensors, aswell as how to use certain isolation and wiring
techniques to minimize or eliminate significant
noise in data-acquisition systems.

Transducer Electronic Data Sheets

Chapter 12 covers Transducer Electronic Data
Sheets. TEDS is a class of so-called smart sensors
that contain an onboard memory chip. The
chip stores information regarding transducer
calibration, manufacturer information, andmany other data.

Measurement Computing

Find application notes, product descriptions, product data sheets, user's manuals,
and other information on MCC's Web site,www.mccdaq.com.

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1

Chapter 1
INTRODUCTION TO DATA ACQUISITION
AND SIGNAL CONDITIONING

SIGNALS, SENSORS, AND SIGNALCONDITIONING

All industrial processing systems, factories,
machinery, test facilities, and vehicles consistof hardware components and computer software whose behavior follow the laws ofphysics as we understand them. These systems
contain thousands of mechanical and electrical phenomena that are continuously
changing; they are not steady state. The
measurable quantities that represent the characteristics of all systems are called variables.
The proper functioning of a particular systemdepends on certain events in time and theparameters of these variables. Often, we areinterested in the location, magnitude, andspeed of the variables, and we use instruments
to measure them. We assign the variables
units of measure such as volts, pounds, andmiles per hour, to name a few.
Most variables must be measured with adevice that converts the phenomena into
a form that a human can perceive such as avisual display, a transducer for sound, orvibrations to stimulate physical sensations.
The conversion devices are called transducers
or sensors, and they translate the physicalphenomena to electrical signals (or viceversa) to be measured with electronic
instruments. These instruments havetraditionally been ammeters, voltmeters,and various other gages, and the variables
can be observed in real time. But anincreasing need to record and preserve these
phenomena and analyze them at a later time

Measurement Computing(508) 946-5100

forced engineers to develop data recorders
and data acquisition systems.
Variables may be classified in many ways, but
generally, most experts prefer two classifications:by characteristic and by type of measurement
signal. Variables classified by characteristic
include thermal, radiation, force, rate, quantity,
time, geometric, physical properties, chemicalcomposition, and electrical. Those classified bymeasurement signal include motion, force, electrical, and time-modulated. Measurement signals
for variables often are hard to differentiate fromthe measuring system. Four factors require close
consideration for measurement signals andsystems: the types of transducers available forconverting variables to measurement signals,
transmission characteristics, data acquisition
system input matching, and transducers available to convert from one type of measurement
signal to another measurement signal.

DATA ACQUISITION SYSTEMS

Data acquisition systems have evolved overtime from electromechanical recorders
containing typically from one to four channels
to all-electronic systems capable of measuring
hundreds of variables simultaneously. Earlysystems used paper charts and rolls or magnetic
tape to permanently record the signals, butsince the advent of computers, particularly
personal computers, the amount of data andthe speed with which they could be collected
increased dramatically. However, many of theclassical data-collection systems still exist andare used regularly.

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2

CHAPTER 1 Introduction to Data Acquisition and Signal Conditioning

PC–BASED DATA ACQUISITION
EQUIPMENT

Early, expensive mainframe computers wereused extensively for gathering multiple channels of data, primarily in large industrial orscientific applications. They were seldom usedin small projects because of their relatively
high cost. But the introduction of small rackmounted minicomputers that developed in
the 1960’s and later desktop personal-typecomputers that housed microprocessors andproliferated in the 1970’s justified their use forsmaller projects. Soon, data acquisition plug-in
cards (as well as hundreds of other types ofplug-in cards) for these small computers werea common means to collect and record data ofall types.
Plug-in cards for computers did not always
perform to the user’s expectations, however.
Internal noise from rotating devices such as
drives and electromagnetic and electrostatic
noise from the computer’s internal bus structureoften interfered with the measured variable,
particularly in data acquisition cards. Isolationand shielding have helped to solve the problem
in most cases, but many data acquisition manufacturers also provide signal conditioning and

Measurement Computing(508) 946-5100

signal processing circuits in small, stand-alone,shielded enclosures. The separate box provides
isolation by distance, expansion for hundreds ofchannels, and portability with laptop computersthat desktop personal computers with plug-in
cards don’t possess.
All PC-based data acquisition systems will record
extremely accurate, repeatable, reliable, anderror-free data provided they are connected andoperated according to the manufacturer’s recommended practices. These practices include
selecting the correct sensors for the application,
the proper wire and shielded cable; capturing
the signals in proper magnitude, range, andfrequency; and paying close attention to
grounding and shielding – particularly eliminating ground loops. Additional items include
choosing the correct impedance and using
doubled-ended (differential) inputs instead of
single-ended where possible. The environment
should also be considered, especially for extremesof ambient temperature, shock, and vibration.
And herein lies the major goal of this publication – to inform users of the most needed recommended practices based upon a fundamental
knowledge of the internal workings of dataacquisition system instrumentation.

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3

Chapter 2
ANALOG-TO-DIGITAL CONVERSIONADC TYPES
Analog-to-Digital Converters (ADCs) transform an analog voltage to a binary number (aseries of 1’s and 0’s), and then eventually to adigital number (base 10) for reading on ameter, monitor, or chart. The number ofbinary digits (bits) that represents the digital
number determines the ADC resolution.
However, the digital number is only an approximation of the true value of the analog voltage
at a particular instant because the voltage canonly be represented (digitally) in discrete
steps. How closely the digital number approximates the analog value also depends on the
ADC resolution.
A mathematical relationship conveniently
shows how the number of bits an ADC handles determines its specific theoretical resolution: An n-bit ADC has a resolution of one
part in 2n. For example, a 12-bit ADC has aresolution of one part in 4,096, where 212 =4,096. Thus, a 12-bit ADC with a maximum
input of 10 Vdc can resolve the measurement
into 10 Vdc/4096 = 0.00244 Vdc = 2.44 mV.
Similarly, for the same 0 to 10 Vdc range, a16-bit ADC resolution is 10/216 = 10/65,536= 0.153 mV. The resolution is usually specified with respect to the full-range reading of
the ADC, not with respect to the measured
value at any particular instant.

Successive-Approximation ADCs
A successive-approximation converter, Figure
2.01 (2.02)
2.01, is composed of aFigure
digital-to-analog
converter (DAC), a single comparator, andMeasurement Computing(508) 946-5100

some control logic and registers. When theanalog voltage to be measured is present at the
input to the comparator, the system control
logic initially sets all bits to zero. Then the
DAC’s most significant bit (MSB) is set to 1,
which forces the DAC output to 1/2 of fullscale (in the case of a 10 V full-scale system,
the DAC outputs 5.0 V). The comparator thencompares the analog output of the DAC to the
input signal, and if the DAC output is lower
than the input signal, (the signal is greaterthan 1/2 full scale), the MSB remains set at 1.
If the DAC output is higher than the input
signal, the MSB resets to zero. Next, the second
MSB with a weight of 1/4 of full scale turns on(sets to 1) and forces the output of the DAC to

Successive-Approximation ADC
DACComparator
Vinput–+Controllogic &
registers

Digitaloutputs

Fig. 2.01. Interestingly, this ADC uses a digital-toanalog converter and a comparator. The logic sets
the DAC to zero and starts counting up, setting eachfollowing bit until it reaches the value of the measured input voltage. The conversion is then finishedand the final number is stored in the register.

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CHAPTER 2 Analog-to-Digital Conversion

either 3/4 full scale (if the MSB remained at 1)
or 1/4 full scale (if the MSB reset to zero). The
comparator once more compares the DAC
output to the input signal and the second bit
either remains on (sets to 1) if the DAC output
is lower than the input signal, or resets to zero
if the DAC output is higher than the input
signal. The third MSB is then compared thesame way and the process continues in orderof descending bit weight until the LSB iscompared. At the end of the process, theoutput register contains the digital code
representing the analog input signal.
Successive approximation ADCs are relatively slow because the comparisons run
serially, and the ADC must pause at eachstep to set the DAC and wait for its output
to settle. However, conversion rates easily
can reach over 1 MHz. Also, 12 and 16-bit
successive-approximation ADCs are relatively inexpensive, which accounts fortheir wide use in many PC-based dataacquisition systems.

frequency, and the pulse counter output, inturn, represents the digital voltage.
Voltage-to-frequency converters inherently
have a high noise rejection characteristic,
because the input signal is effectively integrated over the counting interval. Voltage-tofrequency conversion is commonly used toconvert slow and noisy signals. Voltage-tofrequency ADCs are also widely used forremote sensing in noisy environments. The
input voltage is converted to a frequency at
the remote location and the digital pulse train
is transmitted over a pair of wires to thecounter. This eliminates noise that can be
introduced in the transmission lines of ananalog signal over a relatively long distance.

Integrating ADCs: Dual Slope

A number of ADCs use integrating techniques,
which measure the time needed to charge ordischarge a capacitor in order to determine theinput voltage. A widely used technique, called dual-

Dual-Slope ADC Integration andVoltage-to-Frequency ADCs

Discharge Times

I ∝ Vinput
Vcapacitor

Voltage-to-frequency ADCs convert the analog
input voltage to a pulse train with thefrequency proportional to the amplitude of
the input. (See Figure 2.02.) The pulses arecounted over a fixed period to determine theFixed discharge
current

Voltage-to-Frequency ADC
Vin

Voltage-to-frequency
converter
Digitalpulse train

VinputTi
Integration
time

Timingcircuitry

Pulse
counter

Digitaloutputs

Fig. 2.02. Voltage-to-frequency converters rejectnoise well and frequently
are usedmeasuring
Figure2.03 for(2.04)
slow signals or those in noisy environments.

Measurement Computing(508) 946-5100

Td

Vref

=

Ti
Td

Discharge
time

Fig. 2.03. Dual-slope integrating ADCs provide
high-resolution measurements with excellent noise
rejection. They integrate upward from an unknown
voltage and then integrate downward with a known
source voltage. They are more accurate than single
slope ADCs because component errors are washedout during the de-integration period.

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Data Acquisition Handbook

slope integration, is illustrated in Figure 2.03. It
charges a capacitor over a fixed period with acurrent proportional to the input voltage. Then, thetime required to discharge the same capacitor undera constant current determines the value of the input
voltage. The technique is relatively accurate andstable because it depends on the ratio of rise time tofall time, not on the absolute value of the capacitor
or other components whose values change overtemperature and time.
Integrating the ADC input over an interval
reduces the effect of noise pickup at the ac line
frequency when the integration time ismatched to a multiple of the ac period. For
this reason, it is often used in precision digital
multimeters and panel meters. Although
20-bit accuracy is common, it has a relatively
slow conversion rate, such as 60 Hz maximum,and slower for ADCs that integrate overmultiples of the line frequency.

Sigma-Delta ADCs
Figure(2.06)type of inteA sigma-delta ADC
is 2.04
anothergrating ADC. It contains an integrator, a DAC,
a comparator, and a summing junction. (SeeFigure 2.04.) Like the dual-slope ADC, it’soften used in digital multimeters, panel
meters, and data acquisition boards. Sigmadelta converters are relatively inexpensive
primarily because they have a single-bit DAC,
but they can obtain high-resolution measurements using oversampling techniques.
Although the ADC works best with low-bandwidth signals (a few kHz), it typically has
better noise rejection than many others, andusers can set the integration time (albeit
below 100 samples/sec).
Sigma-delta ADCs also require few external
components. They can accept low-level signals
without much input-signal conditioning
circuitry for many applications, and they don’t
require trimming or calibration components
because of the DAC’s architecture. The ADCs
also contain a digital filter, which lets them

Measurement Computing(508) 946-5100

5

Sigma-Delta ADC
Vin

∑

Vs

Integrator

VIO

–

VDAC
VDAC
VDAC

V CO

+Vref

1

–Vref

0

+

DAC

Vout
to digital
VCO filter

VCO

Vref

Fig. 2.04. Integrating converters such as the sigmadelta ADC have both high resolution and exceptional
noise rejection. They work particularly well for lowbandwidth measurements and reject high-frequency
noise as well as 50/60 Hz interference.

work at a high oversampling rate without aseparate anti-aliasing filter at the input. Sigmadelta ADCs come in 16 to 24-bit resolution,and they are economical for most dataacquisition and instrument applications.
The principle of operation can be understoodfrom the diagram. The input voltage Vinsums algebraically with the output voltage of
the DAC, and the integrator adds the summing
point output Vs to a value it stored previously. When the integrator output is equal toor greater than zero, the comparator output
switches to logic one, and when the integrator output is less than zero, the comparator switches to logic zero. The DAC modulates the feedback loop, which continually
adjusts the output of the comparator to equal
the analog input and maintain the integrator
output at zero. The DAC keeps the integrator’s output near the reference voltage level.
Through a series of iterations, the output
signal becomes a one-bit data stream (at ahigh sample rate) that feeds a digital filter.
The digital filter averages the series of logic
ones and zeros, determines the bandwidth

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6

CHAPTER 2 Analog-to-Digital Conversion

and settling time, and outputs multiple-bit
data. The digital low-pass filter then feeds thedecimation filter, which in turn, decreases
the sample rate of the multi-bit data stream
by a factor of two for each stage within thefilter. For example, a seven-stage filter can
provide a sample-rate reduction of 128.

Improved Accuracy
The digital filter shown in Figure 2.05 inherently improves the ADC’s accuracy for ac
signals in two ways. First, when the input
signal varies (sine wave input) and the system
samples the signal at several times the Nyquist
value (refer to page 17), the integrator becomesa low-pass filter for the input signal, and ahigh-pass filter for the quantization noise. The
digital filter’s averaging function then lowers
the noise floor even further, and combinedwith the decimation filter, the data stream
frequency at the output is reduced. For
example, the modulator loop frequency could
be in the MHz region, but the output data
would be in the kHz region. Second, thedigital filter can be notched at 60 Hz to eliminate power line frequency interference.

Sigma-Delta ADC With Digital Filter and
Decimator Stage
Figure 2.06
(2.04)
Single-bit
Multiple-bit
data stream

Vin

SigmaDelta
converter

data stream

Digitallow-pass
filterModulator loop
frequency = 2 MHz

Decimation
stage

Outputdata

Output data
rate = 2 kHz

Fig. 2.05. Sigma-delta ADCs are well suited to highresolution acquisition because they use oversampling
and often combine an analog modulator, a digital
filter, and a decimator stage. The low-pass digital
filter converts the analog modulator output to adigital signal for processing by the decimator.

Measurement Computing(508) 946-5100

The output data rate from the decimation
filter is lower than the initial sample rate but
still meets the Nyquist requirement by saving
certain samples and eliminating others. Aslong as the output data rate is at least two
times the bandwidth of the signal, the decimation factor or ratio M can be any integer value.
For example, if the input is sampled at fs, theoutput data rate can be fs/M without losing
information. This technique provides morestable readings. (Refer to the table in Figure
2.06 for ADC comparisons.)

Table of ADC Attributes
ADC Type

TypicalResolution

TypicalConversion
Rate*/Frequency

Sigma-Delta

16–24 bit

1 sps–128 ksps

Successive Approximation

8–16 bit

10 ksps–2 Msps

Voltage-to-Frequency

8–12 bit

1 Hz–4 MHz**

Integrating

12–24 bit

1 sps–1 ksps*** sps = samples per second** With line cycle rejection

Fig. 2.06. Table of ADC attributes

ACCURACY AND RESOLUTION
Accuracy is one of the most critical factors toconsider when specifying an ADC for test andmeasurement applications. Unfortunately, it’soften confused with resolution, and althoughrelated, they are distinctly different. Both topics are discussed in this section in some detail,as well as their relationship to calibration, linearity, missing codes, and noise.

Accuracy vs. Resolution
Every ADC measurement contains a variety ofunavoidable, independent errors that influence
its accuracy. When σi represents each independent error, the total error can be shown as:
σtotal =Σ i σi 2

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7

Data Acquisition Handbook

words, the input analog voltage range ispartitioned into a discrete number of values
that the converter can measure, which is also
the ADC’s resolution. The quantization error inthis case is specified to be no more than half ofthe least significant bit (LSB). For the 12-bit
ADC, the error is ±1.22 mV (0.0122%). Such
ADC errors are typically specified in three ways:
the error in LSBs, the voltage error for a
specified range, and the % of reading error.
Most ADCs are not as accurate as their specified
resolution, however, because other errors
contribute to the overall error such as gain,
linearity, missing codes, and offset. (See Figures
2.07B, C, D, and E, respectively). Nonetheless,the accuracy of a good ADC should approach
its specified resolution. When an ADC

This equation includes a variety of errors suchas sensor anomalies, noise, amplifier gain andoffset, ADC quantization (resolution error),and other factors.

Quantization Error
In a theoretically perfect ADC, any particularanalog voltage measured should be represented
by a unique digital code, accurate to an infinite
number of digits. (See Figure 2.07A.) But in areal ADC, small but finite gaps exist between
one digital number and a consecutive digital
number, and the amount depends on thesmallest quantum value that the ADC canresolve. In the case of the 12-bit converter
covering a 10 Vdc range, for example, thatquantum value is 2.44 mV, the LSB. In otherCommon ADC Errors
A

Ideal

7

7

6

ADC Output

ADC Output

6

B

Gain error

5
4
3
2

5
4
3
2

1

1

0

0

0

2

4

6

Input volts

8

10

Missing code

7

C

5
4
3
2
1

0

2

4

6

8

Input volts

D
7

6

0

10

0

2

6

8

4

6

Input volts

8

10

E

Offset error

ADC Output

ADC Output

6

5
4
3
2

5
4
3
2
1

1
0

Linearity error

6

ADC Output

7

0

2

4

6

Input volts

8

10

0

0

2

4

Input volts

10

Fig. 2.07. The straight line in each graph represents the analog input voltage and the perfect output voltage reading from an ADC with infinite resolution. The step function in Graph A shows the ideal response for a 3-bit ADC.
Graphs B, C, D, and E show the effect on ADC output from the various identified errors.

Figure 2. 07 (2.05)
Measurement Computing(508) 946-5100

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CHAPTER 2 Analog-to-Digital Conversion

manufacturer provides calibration procedures,
offset and gain errors usually can be reduced tonegligible levels, however, linearity andmissing-code errors are more difficult or
impossible to reduce.

ADC Accuracy vs. System Accuracy
Calibration
ADCs may be calibrated with hardware, software, or a combination of the two. Calibrationin this case means adjusting the gain andoffset of an ADC channel to obtain the specified input-to-output transfer function. In a
hardware configuration, for example, theinstrumentation amplifier driving the ADC
has its offset and gain adjusted with trim
pots, and changing the ADC’s reference
voltage changes its gain. In hardware/software calibrations, the software instructs DACs
to null offsets and set full-scale voltages.
Lastly, in a software calibration, correction
factors are stored in nonvolatile memory inthe data acquisition system or in the computerand are used to calculate the correct digitalvalue based on the readings from the ADC.
ADCs are factory calibrated before being
shipped, but time and operating temperature
can change the settings. ADCs need to be recalibrated usually after six months to a year, andpossibly more often for ADCs with resolutionsof 16 bits or more. Calibration proceduresvary, but all usually require a stable reference
source and an indicating meter of (at least
three times) greater accuracy than the device
being calibrated. Offset is typically set to zerowith zero input, and the gain is set to full scalewith the precise, full-scale voltage applied tothe input.
In many measurements, the input voltage
represents only the physical quantity under
test. Consequently, system accuracy might be
improved if the complete measurement
system is calibrated rather than its individual
parts. For example, consider a load cell with

Measurement Computing(508) 946-5100

output specified under a given load and excitation voltage. Calibrating individual partsmeans that the ADC, load cell, and excitation
source accuracy tolerances are all added
together. With this approach, the error sourcesof each part are added together and generate
a total error that is greater than the error thatcan be achieved simply by calibrating thesystem with a known precision load andobtaining a direct relationship between the
input load and ADC output.

Linearity
When the input voltage and the ADC output
readings deviate from the diagonal line (representing infinite resolution) more than the ideal
step function shown in Figure 2.07A, the ADC
error is nearly impossible to eliminate by calibration. The diagonal line represents an ideal,
infinite-resolution relationship between input
and output. This type of ADC error is called anonlinearity error. Nonlinearities in a calibrated ADC produce the largest errors near the
middle of the input range. As a rule of thumb,nonlinearity in a good ADC should be one LSBor less.

Missing Codes
A quality ADC should generate an accurate
output for any input voltage within its resolution, that is, it should not skip any successive
digital codes. But some ADCs cannot produce
an accurate digital output for a specific analog
input. Figure 2.07D, for example, shows that a
particular 3-bit ADC does not provide an
output representing the number four for any
input voltage. This type of error affects boththe accuracy and the resolution of the ADC.

Noise
The cost of an ADC is usually proportional toits accuracy, number of bits, and stability. But
even the most expensive ADC can compromise accuracy when excessive electrical noiseinterferes with the measured signal, whether
that signal is in millivolts or much larger.

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Data Acquisition Handbook

For example, many ADCs that reside on cards
and plug into a PC expansion bus can encounter
excessive electrical noise that seriously affects
their accuracy, repeatability, and stability. But
an ADC does not have to be connected directly
to the bus within the computer. An ADC
mounted in an external enclosure often solves
the problem. It can communicate with thecomputer over an IEEE 488 bus, Ethernet, serial
port, or parallel port.
When there is no choice but to locate an ADC
inside the computer, however, check its noiselevel. Connecting the ADC’s input terminal
to the signal common terminal should
produce an output of zero volts. If it stillreads a value when shorted, the noise is being
generated on the circuit card and will interfere with the desired input signal. More critical diagnostics are necessary when using an
external power supply because noise also canarise from both the power supply and the
input leads.

Noise Reduction and Measurement Accuracy
One technique for reducing noise andensuring measurement accuracy is to eliminate
ground loops, that is, current flowing in the
ground connection between different devices.
Ground loops often occur when two or more
devices in a system, such as a measurement
instrument and a transducer, are connected to
ground terminals at different physicallocations. Slight differences in the actual
potential of each ground point generate acurrent flow from one device to the other.
This current, which often flows through the
low potential lead of a pair of measurement
wires generates a voltage drop that appears as
noise and measurement inaccuracy at thesignal conditioner or ADC input. When atleast one device can be isolated, such as thetransducer, then the offending ground path isopen, no current flows, and the noise or
inaccuracy is eliminated. Optical isolators,
special transformers, and differential input

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9

operational amplifiers at the signal conditioner
or ADC input can provide this isolation.

ADC Noise Histograms
ADC manufacturers frequently verify theirdevice’s accuracy (effect of non-linearities) byrunning a code density test. They apply ahighly accurate sine wave signal (precisionamplitude and frequency) to the device andusing a histogram for analysis, generate a
distribution of digital codes at the output of
the converter. A perfect ADC would produce
only one vertical bar in the histogram for the
specified input frequency and amplitude
because it measured only one value for every
sample. But because of the ADC’s inherent
non-linearities, it produces a distribution ofbars on either side representing digital words
sorted into different code bins. Each bin islabeled for a single digital output code and it
contains the count of its occurrence, or thenumber of times that code showed up in the
output. (See Figure 2.08.)
When n represents the ADC’s bit resolution,
2n bins are required. The width of each codebin should be FSR/2n where FSR is the fullscale range of the ADC. The probability
density function may be determined from thisdata. A large number of samples must be
taken, depending on the ADC’s bit size, forthe histogram test to be meaningful. The
more bits the ADC contains, the higher thenumber of samples required, which could be
as much as 500,000 samples.

ENOB: Effective Number of Bits
Although an ADC’s accuracy is crucial to adata acquisition system’s accuracy, it’s not the
final word. A widely used and practical way ofdetermining overall measurement accuracy isaccomplished with what’s called an Effective
Number of Bits (ENOB) test. The ENOB mayvery well demonstrate that the actual systemmeasurement accuracy is something less than

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CHAPTER 2 Analog-to-Digital Conversion

Histogram
Bin with most samples
represents the input signalof 2.50 VDCHistogram shape
approximates a Gaussian
distribution caused bywhite noise mixed
with dc input voltage
Number of samples

Bins with samples
caused by noise

0

1019

1020

1021

1022

1023

1024

1025

1026

1027

1028

1029

4095

Total number of bins = 4096
FSR

Fig. 2.08. The histogram illustrates how 12-bit ADC samples in a set were distributed among the various codes
for a 2.5 V measurement in a FSR (full-scale range) of 10 V. Most codes intended for the 1024 bin representing
2.5 V actually ended up there, but others fell under a gaussian distribution due to white noise content.

the ADC bit accuracy specifications. For
example, an ADC may be specified as a 16-bit
device,but2.08
the results of a specific type of
Figurestandard test may show that its performance isactually closer to an ideal 13-bit system.
However, 13 bits may be more than adequate
for the application.
The ENOB test takes into account all thecircuitry from input terminals to the data
output, which includes the effects of the
ADC, multiplexer, and other analog anddigital circuits on the measurement accuracy.
It also includes the signal-to-noise ratio, SNR,or the effect of any noise signals induced into
the system from any source.

The ENOB Test
The ENOB test evaluates the data acquisition
system as it performs in a real-world application
when used with the manufacturers recommended
cables, connectors, and connections. It considers
the front end of the data acquisition system: theMeasurement Computing(508) 946-5100

ADC, multiplexer, programmable gain amplifier,and sample-and-hold amplifiers. All of thesecircuits affect the digitized output. Any
non-linearities, noise, distortion, and otheranomalies that sneak into the front end can
reduce the system’s accuracy, and it’s not enoughto test only one channel in a multi-channel
system. Some errors creep up from the effect of
one channel on another through cross talk.
To measure the ENOB, set up a precision sinewave signal generator and connect it’s output
to the input of one analog input channel. The
generator itself should produce little noiseand distortion. Set the signal generator output
amplitude to just under the maximum
specified input range of the board. Set thegenerator to the maximum frequency that thesystem is specified to measure. Next, groundthe input terminals of the adjacent channel.
Run the system at its maximum rated speed.
Sample the test signal and then the grounded
input. Capture 1024 samples on each input

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Data Acquisition Handbook

and run the samples through an FFT algorithm
to compute the ENOB.
The test measures the effects of slewing,
harmonic distortion, analog circuits, ADC
accuracy, noise pickup, channel cross talk,integral and differential non-linearity, andoffset between channels. Although ENOB
provides a good benchmark of the systems
capability and accuracy, it is not a specification; it is not a replacement for SNR and othererror specifications provided by the manufacturer. However, systems may be comparedwith ENOB tests when all are measured under
the same set of conditions.

ADC OUTPUT AVERAGING BENEFITS
Improved Accuracy

A paradox that arises from averaging theoutput of an integrating-type ADC is that themeasurement system theoretically can obtainhigher accuracy for a signal embedded in noise
than a signal free of noise. How this can betrue comes from the way a signal ismathematically averaged. For example, with asingle dc signal, averaging the output always
provides the same result with no apparentchange in accuracy (not considering the affectsof calibration). But for a varying input signal,such as a sine wave, a large number of samples
yield a Gaussian distribution which can be
accurately defined with a more precisely
established peak for the wave. But the samples
must not all cluster around a specific portionof the sampled wave. To get a true distribution,

Measurement Computing(508) 946-5100

11

the ADC must sample at a slower rate than thefluctuations and be out of synch with them.
This technique finds a general average, not a
local average. Thus, in this way signal averaging
increases the system’s measurement resolution.

More Stable Readings
Some systems actually introduce a random
noise signal called dither into an otherwise
clean ADC input to take advantage of the averaging function for increasing accuracy andsignal stability. The technique also enables an
ADC with a smaller number of bits to obtain
the resolution of an ADC with more bits
without losing accuracy. Each signal sweep
must capture a different random value at eachpoint in time. Then the theoretical ADC average
from this distribution will remain at zero over a
sufficiently large sample window. For example,
if 16 values are averaged, then it has 16 times
more possible values than the direct non-averaged output. This technique increases the effective ADC resolution by 4 bits. The noise makes
it work.
ADC signal averaging is so important atechnique that it is used on digital recordings.
In the early days of digital audio recording
development, systems lacked ADC output
averaging. As a result, a musical note would
decay into an annoying buzz because not allbits in the ADC were enabled. The output
waveforms were distorted and the ear could
not filter it out, but ADC signal averaging
totally eliminated the problem.

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12

CHAPTER 2 Analog-to-Digital Conversion

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13

Chapter 3
MULTIPLEXING AND SAMPLING THEORYTHE ECONOMY OF MULTIPLEXING
Sampled-Data Systems

An ideal data acquisition system uses asingle ADC for each measurement channel.
In this way, all data are captured in paralleland events in each channel can be comparedin real time. But using a multiplexer, Figure
3.01, that switches among the inputs ofmultiple channels and drives a single ADC
can substantially reduce the cost of asystem. This approach is used in so-calledsampled-data systems. The higher the
sample rate, the closer the system mimics
the ideal data acquisition system. But only
a few specialized data acquisition systems
require sample rates of extraordinary speed.
Most applications can cope with the more

Multiplexers
AnaloginputsMux

Gaincontrol lines
ADC

Fig. 3.01. The multiplexer (MUX) is a fast switchthat sequentially scans numerous input-signal channels and directs them in a preprogrammed manner toa single ADC for digitizing. This approach saves thecost of using an ADC for each channel.

Measurement Computing

Solid State Switches vs. Relays

A multiplexer is an array of solid-state switches
or electromechanical relays connected toseveral input channels. Although both
approaches are used in a wide variety of applications, neither one is perfect; each typecomes with various advantages and disadvantages. Electromechanical relays, for example,
are relatively slow, about 1,000 samples/sec orless for the fastest reed relays, but they canhandle large input voltages, and some canisolate voltages of several kV. A relay’s size andcontact type determine its current carrying
capacity. For instance, laboratory instrument
relays typically switch up to 3 A, while industrial applications use larger relays to switch
higher currents, often 5 to 10 A.
Solid-state switches, on the other hand, aremuch faster than relays and can reach sampling
rates of several MHz. However, these devices
can’t handle inputs higher than 25 V, and they
are not well suited for isolated applications.
Moreover, solid-state devices are typically limited
to handling currents of only one mA or less.

PGA

Mux
control lines

modest sample rates typically offered bymainstream data acquisition systems.

(508) 946-5100

Another characteristic that varies between
mechanical relays and solid-state switches iscalled ON resistance. An ideal mechanical switch
or relay contact pair has zero ON resistance. But
real devices such as common reed-relay contacts
are 0.010 Ω or less, a quality analog switch canbe 10 to 100 Ω, and an analog multiplexer canbe 100 to 2,500 Ω per channel. The ON resis-

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14

CHAPTER 3 Multiplexing and Sampling Theory

tance adds directly to the signal source impedance and can affect the system’s measurement
accuracy if not compensated.
Analog switching devices have another undesirable characteristic called charge injection.
This means that a small portion of input-gatedrive voltage is coupled to the analog input
signal and manifests as a spike in the output
signal. This glitch produces measurement
errors and can be seen riding on the input
signal when the source impedance is too high.
A compensating circuit can minimize theeffects of charge injection, but the most effective method is to keep source impedance as
low as possible to prevent it from developingin the first place.
Channel-to-channel cross talk is another nonideal characteristic of analog switching
networks, especially integrated circuit multiplexers. Cross talk develops when the voltage
applied to any one channel affects the accuracy of the reading in another channel.
Conditions are optimum for cross talk when
signals of relatively high frequency and highmagnitude such as 4 to 5 V signals areconnected to one channel while 100 mV
signals are connected to an adjacent channel.
High frequency multiplexing also exacerbates
cross talk because the signals couple through asmall capacitance between switch channels.
Low source impedance minimizes the crosstalk and eliminates the charge injection.

Speed

Multiplexing reduces the rate at which data
can be acquired from an individual channel
because of the time-sharing strategy between
channels. For example, an ADC that cansample a single channel at 100 kHz is limited
to a 12.5 kHz/channel sampling rate whenmeasuring eight channels.
Unfortunately, multiplexing can introduce
yet other problems. For instance, the multiplexer’s high source impedance can combine

Measurement Computing(508) 946-5100

with stray capacitance to increase settling
time and generate cross talk between channels. Multiplexer impedance itself also candegrade signals. A solid-state multiplexer with
an impedance of tens or hundreds of ohms isworse than a relay with a typical resistance of
0.010 Ω or less.
In spite of these negative issues, the advantages of multiplexing outweigh its disadvantages, and it has become a widely used technique to minimize cost without compromising
performance. Because the measurement errors
are known and specified, they can be compensated at each stage of the data acquisition
system to ensure high accuracy at the output.

Sequence vs. Software-Selectable Ranges
Most data acquisition systems accommodate avariety of input ranges, although the manner inwhich they do so varies considerably. Some
data acquisition systems allow the input range
to be switched or jumper selected on the circuitboard. Others provide software-selectable gain.
This is more convenient, but a distinction
should be made between data acquisition
systems whose channels must all have the samegain and other systems that can sequentiallyselect the input range for each channel. The
more useful system accepts different input
ranges on different channels, especially whenmeasuring signals from different transducers.
For example, thermocouples and strain gages
require input ranges in tens of mV and use
special signal conditioners, while other sensors
might output several volts.
A data acquisition system with a softwareselectable range can measure different ranges
on different channels (but at a relatively slowrate) with a command to change the gain
between samples. But the technique has two
problems. First, it is relatively slow. That is,issuing a software command to change thegain of a programmable-gain amplifier (PGA)
can take tens or hundreds of ms, lowering theinfo@mccdaq.com

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Data Acquisition Handbook

15

drastic reduction in the scan rate for expansion
channels, a major limitation encountered withmany plug-in data acquisition boards.

system’s sample rate to several Hz. Second, thespeed of this sequence is often indeterminate
due to variations in PC instruction cycle times.
Cycling through the sequence continuously
generates samples with an uneven (andunknown) spacing in time. This complicatestime-series analysis and makes FFT analysis
impossible because its algorithm requires
evenly spaced samples.

All channels are scanned, including expansion
channels, at 100 kHz, (10 µs per channel). (SeeFigure 3.03.) Digital inputs also can be scanned
using the same scan sequence employed foranalog inputs, enabling the time correlation of
acquired digital data to acquired analog data.
Such systems permit each scan group (containing up to 512 channel/gain combinations) tobe repeated immediately or programmed tointervals of up to 12 hours. Within each scangroup, consecutive channels are measured at a
fixed 10 µs per channel rate.

A better implementation hosts a sequencer
that sets the maximum acquisition rate andcontrols both channel selection and associatedamplifier gain at random. For example, one
widely used data acquisition system runningat 100 kHz and 1 MHz uses software selectablechannel gain and sequencing. (See Figure
3.02.) The 100 kHz system provides a
512-location scan sequencer that lets operators
use software to select each channel and its
input amplifier gain for both the built-in andexpansion channels. Each scan group can be
repeated immediately or at programmableintervals. The sequencer circuitry overcomes a

FUNDAMENTAL CONCEPTS
Sampling Rates

When ADCs convert an analog voltage to adigital representation, they sample the value ofthe measured variable several times per second.
Steady or slowly changing dc voltages might
require sampling rates of only several Hz, but

Scan Sequencer
All channels within a scan group aremeasured at a fixed scan rate

Scan groupt

Programmable
Constant
scan rate
Channel1 #2
Gain2 x1
Unipolar or Bipolar3 Uni

#4
x8
Uni

#7
x8
Bi

#2
x2
Uni

D4

#18 5
x100
Bi

#19
x10
Bi

#16
x1000
Uni

t

1. Unipolar or bipolar operation can be programmed for each channel dynamically by the sequencer
2. Gain can be programmed for each channel dynamically by the sequencer
3. Channels can be sampled dynamically by the sequencer
4. Expansion channels are sampled at the same rate as on-board channels

Fig. 3.02. This is an example of a 512-location scan sequencer operating in a 100 kHz data acquisition system.
The sequencer can program each channel dynamically for unipolar or bipolar operation. The sequencer also canprogram the gain for each channel on the fly and change the order in which the channels are scanned.

Figure 3.02
Measurement Computing(508) 946-5100

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16

CHAPTER 3 Multiplexing and Sampling Theory

Multiplexer With
Sequence-Selectable Gain
Analoginputs16
to1
Mux

Expansionchanneladdressing

4

2
Expansion
gain addressing

x1, x2, x4, & x8
PGA

Hardwaresequencer

BufferTo dataacquisition
systemPGA
BufferMux

BufferADC

CLK
Triggering
circuitryFig. 3.03. A multiplexer with sequence-selectable
gain lets the data system operate at its maximum
acquisition rate.

Figure 3.04

measuring various ac and sine waves is different.
Enough samples/sec must be made to ensurethat the wave being measured in the continuous-time domain can be faithfully reproduced
in both the continuous-time domain and thediscrete-time domain.

Source Impedance

Most signal sources have impedances of lessthan 1.5 kΩ, so such a maximum source impedance is usually not a problem. However, fastermultiplexer rates require lower source impedances. For example, a 1 MHz multiplexer in a
12-bit system requires a source impedance lessthan 1.0 kΩ. When the source impedanceexceeds this value, buffering is necessary to
improve accuracy. A buffer is an amplifier witha high input impedance and extremely low
output impedance. (See Figure 3.04.) A buffer
on each channel located between the transducer and the multiplexer ensures higher accuracies by preventing the multiplexer’s stray
capacitance from discharging through the
impedance of the transducer.

Measurement Computing

Inputs

Buffer2

4

Buffers(508) 946-5100

Fig. 3.04. Buffering signals ahead of the multiplexer
increases accuracy, especially with high-impedancesources or fast multiplexers

Sample-and-Hold ADCs
Time Skew

A multiplexed ADC measurement introduces
a time skew among channels, because eachchannel is sampled at a different time. Someapplications can’t tolerate this effect. But asample and hold circuit placed on each input
ahead of the multiplexer remedies time-skew
problems. In a simultaneous sample and holdcircuit (SS&H) each channel is equipped witha buffer that samples the signal at the beginning of the scan sequence. The buffer output
holds the sampled value while the multiplexer switches through all channels, and the
ADC digitizes the frozen signals. In a good
simultaneous sample and hold implementation, all channels are sampled within 100 ns
of each other.
Figure 3.05 shows a common scheme for
SS&H. Each input signal passes through an
instrumentation amplifier (IA), a low-passfilter, and into a sample-and-hold buffer (S/H).
When the sample enable line goes high, each
S/H samples its input signal and holds it while
the multiplexer switches through the readings. This scheme ensures that all the samples

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Data Acquisition Handbook

Aliasing and Fourier Transforms

Ch 2
Ch 3
Ch 4

IA

LPF

S/H

IA

LPF

S/H

IA

LPF

S/H

IA

LPF

S/H

Mux

To dataacquisition
systemChanneladdresslines

Sample enable

Switched bias resistors(per channel)

Fig. 3.05. A four-channel multiplexer with low-passfilters and simultaneous sample-and-hold circuits
ensures that all the samples are taken within a few
ns of each other.

are taken within 50 ns of each other, evenwith up to 256 simultaneous channels
connected to a single instrument.

Nyquist Theorem

Transforming a signal from the time domain to
the frequency domain requires the applicationof the Nyquist theorem. The Nyquist sampling
theorem states that if a signal only contains
frequencies less than cutoff frequency, fc, all
the information in the signal can be captured
by sampling it at a minimum frequency of 2fc.
This means that capturing a signal with amaximum frequency component of fmax
requires that it must be sampled at 2fmax orhigher. However, common practice dictatesthat while working in the frequency domain,
Figure 3.06 (2.09)
the sampling rate must be set more than twiceand preferably between five and ten times thesignal’s highest frequency component.
Waveforms viewed in the time domain areusually sampled 10 times the frequency being
measured to faithfully reproduce the original
signal and retain accuracy of the signal’s
highest frequency components.

Measurement Computing(508) 946-5100

When input signals are sampled at less thanthe Nyquist rate, ambiguous signals that aremuch lower in frequency than the signal
being sampled can appear in the time domain.
This phenomenon is called aliasing. For
example, Figure 3.06 shows a 1 kHz sine wave
sampled at 800 Hz. The reconstructed or transformed frequency of the sampled wave ismuch too low and not a true representation of
the original. If the 1 kHz signal were sampled
at 1,333 Hz, a 333 Hz alias signal would
appear. Figure 3.07, on the other hand, shows
the signals when sampling the same 1 kHz
wave at more than twice the input frequency
or 5 kHz. The sampled wave now appears
closer to the correct frequency.

Inadequate Nyquist Sampling Rates
1
0.8
0.6

Amplitude (V)Sample and Hold
Ch 1

17

0.4
0.2
0
-0.2
-0.4
-0.6
-0.8
-1

0

100

200

300

400

500

600

Frequency (Hz)Original Sine Wave
Sampled less than 1 time per cycle

Fig. 3.06. When sampling an ac signal at less than
two times the Nyquist frequency, the original waveform cannot be reproduced faithfully.

Conversely, input frequencies of half or moreof the sampling rate will also generate aliases.
To prevent these aliases, a low-pass, antialiasing filter is used to remove all components of these input signals. The filter isusually an analog circuit placed between thesignal input terminals and the ADC. Although
the filter eliminates the aliases, it also prevents
any other signals from passing through thatinfo@mccdaq.com

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CHAPTER 3 Multiplexing and Sampling Theory

Acceptable Nyquist Sampling Rates
1
0.8

Amplitude (V)

0.6
0.4
0.2
0
-0.2
-0.4
-0.6
-0.8
-1

100

200

300

400

500

Frequency (Hz)600

Original Sine Wave
Sampled about 4 times per cycle

Fig. 3.07. When ac inputs are sampled more than
twice the Nyquist frequency of the sine wave, thefrequency content of the signal is preserved, and all
the Fourier components of the periodic waveform
are recovered.

are above the stop band of the filter, whether
they were wanted or not. In other words,
when selecting a data acquisition system,
make certain that the per channel sampling
frequency is more that twice the highest
frequency intended to be measured.

Fourier Transforms
1.4
1.2
1
0.8
0.6
0.4

Fourier Transforms

0.2

1

0

0.9
0.8

0

500

1000

Frequency [Hz]

1500

2000

500 Hz square wave
4 kHz sample rate
2 kHz low-pass filter

0.7

Volts

Another example of aliasing is shown in Figure
3.08 for a square wave after passing through a
Fourier transform. A Fourier transform is afrequency spectrum display of the sampled data.
It shows how much energy at a given frequency
is in a particular signal. For the purpose of illustration, assume that the example processes only
frequencies under 2 kHz. Ideally, a Fourier transform of a 500 Hz square wave contains one peakat 500 Hz, the fundamental frequency, andanother at 1,500 Hz, the third harmonic, whichis 1/3 the height of the fundamental. Figure
3.08, however, shows how higher frequency
peaks are aliased into the Fourier transform’s
low-frequency range. The low-pass filter with
cutoff at 2 kHz shown in Figure 3.09 removes
most of the aliased peaks.

Volts

18

0.6
0.5
0.4

Fig. 3.09. The Fourier Transform of a 500 Hz
square wave is sampled at 4 kHz with a low-passfilter cutoff at 2 kHz.

Figure 3.09 (2.11)

0.3
0.2
0.1
0

0

500

1000

Frequency [Hz]500 Hz square wave
4 kHz sample rate
No filters

1500

2000

Fig. 3.08. The Fourier transform of a 500 Hz squarewave is sampled at 4 kHz with no filtering.

Measurement Computing(508) 946-5100

When the sampling rate increases to four times
the highest frequency wanted, the Fouriertransform in the range of interest looks evenbetter. Although a small peak remains at 1 kHz,
it is most likely the result of an imperfectsquare wave rather than an effect of aliasing.
(See Figure 3.10).

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Data Acquisition Handbook

interval produces sampled data that are only
approximations. Consequently, the resolutionof the Fourier transform is limited to approximately 1/T where T is the finite time interval
over which the measurement was made.
Fourier transform resolution can only be
improved by sampling for a longer interval.

1.6
1.4

Volts

1.2
1
0.8
0.6
0.4
0.2
0

500

1000

Frequency [Hz]

1500

2000

500 Hz square wave
8 kHz sample rate
2 kHz low-pass filter

Fig. 3.10. The Fourier Transform of a 500 Hz
square wave is sampled at 8 kHz with a low-passfilter cutoff at 2 kHz.

Discrete Fourier TransformWhen ac signals pass through a time-invariant,linear system, their amplitude and phasecomponents can change but their frequencies
remain intact. This is the process that occurs
when the continuous time domain ac signal
passes through the ADC to the discrete time
domain. Sometimes, more useful information
can be obtained from the sampled data by
analyzing them in the discrete time domain
with a Fourier Series rather than reconstructing
the original signal in the time domain.
The sampled data pass through a Fourier transform function to cull out the fundamental andharmonic frequency information. The amplitude of the signal is displayed in the verticalaxis, and the frequencies measured are plottedon the horizontal axis.

Windowing

Real-time measurements are taken over finite
time intervals. In contrast, Fourier transforms
are defined over infinite time intervals, so
Figure 3.11 (2.13)
limiting the transform to a discrete time

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A finite time interval used for a Fourier
Transform also generates spurious oscillations in the transform’s display as shown in
Figure 3.11. From a mathematical viewpoint,
the signal that’s instantaneously turned onat the beginning of the measurement andthen suddenly turned off at the end of themeasurement produces the spurious oscillations. These spurious oscillations are usuallyeliminated with a function called windowing,that is, multiplying the sampled data by aweighting function. A window function thatrises gradually from zero decreases the
spurious oscillations at the expense of aslight loss in triggering resolution. Many
possible window functions may be used,
including Hanning, Hamming, Blackman,
Rectangular, and Bartlett. (See Figures 3.12,
3.13, and 3.14.)

Fourier Transforms
1.2
1

Amplitude

Fourier Transforms

0

19

0.8
0.6
0.4
0.2
0
450 460 470 480 490 500 510 520 530 540 550

Frequency [Hz]Fourier transform of a sine wave using a window function
Fourier transform of a sine wave with no window function

Fig. 3.11. A Fourier Transform is shown with andwithout a window function.

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CHAPTER 3 Multiplexing and Sampling Theory

Measurement With
Window Function

1

1

0.5

0.5

0

-0.5

Amplitude2
1.5

Amplitude

2
1.5

-1
-1.5

-2

-2

Fast Fourier Transforms

1.5
1

0.5
0

Fig. 3.13. In this case, the frequency information is preserved
when multiplied by a windowing
function. It minimizes the effect ofirregularities at the beginning andend of the sample segment.

HammingHanning
Blackman

Fig. 3.14. The three most common types of window functions
include Hamming, Hanning, andBlackman.

Figure 3.14 (2.16)

The Fast Fourier Transform (FFT) is so common
today that FFT has become an imprecise
synonym for Fourier transforms in general.
The FFT is a digital algorithm for computing
Fourier transforms of data discretely sampled
at a constant interval. The FFT’s simplest
implementation requires 2n samples. Other
implementations accept other special numbersof samples. If the data set to be transformed
has a different number of samples thanrequired by the FFT algorithm, it is often
padded with zeros to meet the required
number. Sometimes the results are inaccurate,
but most often they are tolerable.

Standard Fourier Transforms

A Standard Fourier Transform (SFT) can be used
in applications where the number of samples
cannot be arranged to fall on one of the special
numbers required by an FFT, or where it cannot
tolerate the inaccuracies introduced by padding
with zeros. The SFT is also suitable where thedata are not sampled at evenly spaced intervals
or where sample points are missing. Finally, theSFT can be used to provide more closely spaced
points in the frequency domain than can be

Measurement Computing

2

-0.5

-1

Fig. 3.12. The frequency information obtained from a measurement
can be incorrect when the waveform contains abrupt beginning or
endingpoints.
Figure3.13 (2.15)

2.5

0

-1.5

Common Window
Function

Amplitude

Measurement Without
Window Function(508) 946-5100

obtained with an FFT. In an FFT, adjacent points
are separated by 1/T, the inverse of the timeinterval over which the measurement was made.
Many standard numerical integration techniques exist for computing SFTs from sampled
data. Any other technique selected for the
problem at hand probably will be much slower
than an FFT of a similar number of points. Thisis becoming less of an issue, however, as the
speed of modern computers increases.

Digital vs. Analog Filtering

Digital filtering requires three steps. First, thedigital signal is subjected to a Fourier transform.
Then the signal amplitude in the frequency
domain is multiplied by the desired frequency
response. Finally the transform signal is inverse
Fourier transformed back into the time domain.
Figure 3.15 shows the effect of digital filtering ona noisy signal. The solid line represents the unfiltered signal while the two dashed lines show theaffect of different digital filters. Digital filters
have the advantage of being easily tailored toany frequency response without introducing
phase error. However, one disadvantage is that adigital filter cannot be used for anti-aliasing.

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Data Acquisition Handbook

Digital Filtering
1.546
1.544

Volts

1.542
1.54
1.538
1.536
1.534

0

0.05

0.1

0.15

Time [seconds]

0.2

0.25

Unfiltered signal
50 Hz low-pass digital filter
5 Hz low-pass digital filter

Fig. 3.15. The solid line is the noisy, unfilteredsignal, and the two dashed lines show the same
signal at the output of the 5 and 50 Hz low-passdigital filters.

In contrast to digital filters, analog filters canbe used for anti-aliasing. But changing thefrequency response curves is more difficult
because all analog filters introduce somephase error.

Settling Time
Source impedance and stray capacitanceinfluence the multiplexer input settling time.
Their effect can be predicted with a simple
equation:
EQN. 3.01 Time Constant

EQN. 3.02 Settling Time

Tsample = Tscan – Tsettle
Tsample = 10 µs – 8 µs
Tsample = 2 µs
In a typical 16-bit data acquisition system, theinternal settling time (Tint) may be 6 µs. The
external settling time may then be computed
as follows:
EQN. 3.03 External Settling TimeT ext =

T settle 2 – T int 2

Text = 5.29 µs
For a 16-bit data acquisition system with 100 pF
input capacitance, (Cin) and a multiplexer resistance (Rmux) of 100 Ω, the maximum external
resistance is:

R ext =T ext
(C in )(l n 2 16 ) – R mux

Rext = 4,670 ΩFor example, determine the maximum tolerable
source impedance for a 100 kHz multiplexer.
The time between measurements on adjacentchannels in the scan sequence is 10 µs. During
time T = RC, the voltage error decays by a factorof 2.718, or one time constant, T. But a time

Equation 3.04

interval 10 times longer (10 T) is required toreduce the error to 0.005%. Consequently, a
fixed time of 10 µs between scans (Tscan) and anerror of 0.005% requires that T = 1 µs. But in a
typical multiplexed data acquisition system, thisis insufficient settling time, so the data would
continue to be in error. The relationship can be
explained as follows. Most 100 kHz converter’ssample and hold circuits are set to acquire thesignal at about 80% of the sample window(10 µs) to allow 8 µs of settling time. Subtracting
this from the scan time results in a samplingtime of:

EQN. 3.04 External Resistance

T = RC,
Where:
T = time constant, s
R = source impedance, ΩEquation 3.03
C = stray capacitance, F

Measurement Computing

21(508) 946-5100

The simplified examples above do not include
effects due to multiplexer charge injection or
inductive reactance in the measurement wiring.
In actual practice, the practical upper limit on
source resistance is between 1.5 k and 2 kΩ.

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CHAPTER 3 Multiplexing and Sampling Theory

Measurement Computing(508) 946-5100

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23

Chapter 4
ELECTRICAL MEASUREMENTS

SAFETY CONCERNSIn the United States, electrical potentials
more than 48 V are considered dangerous tohuman life. Although people have survived
electrical shocks exceeding thousands of voltsand lightning strikes carrying millions ofjoules of energy, everyone should treat electricity with caution and respect. Voltage is
not the only concern, however. Currents inthe mA range can stop a normal heart undercertain conditions. For example, NIOSH(National Institute for Occupational Safety
and Health) estimates that as little as 20 mA
at 110 Vac, the common utility voltage, canparalyze respiratory muscles.
All equipment and sources of voltage andcurrent, such as the item under test, should not
be powered when making connections between
sensors, actuators or other components and thedata acquisition system. This is especially critical when using current transformers andmeasuring high voltages and ac current.

MAKING VALID MEASUREMENTS

Accurate measurements depend on accurate
equipment and robust connections. Poor
connections and noise picked up throughunshielded leads and improperly groundedcircuits will certainly yield inaccurate data.
Packaged sensors are typically designed toensure accurate output signals with propercomponent placement, shielding, filtering,and bypassing power and ground I/O connections. Unprotected sensors, however, such as
thermocouples and strain gages often are

Measurement Computing(508) 946-5100

exposed to numerous types of noise signalsthat can couple into the sensors and leads andcontaminate the desired signal. Those who
install such sensors must follow industryaccepted practices to ensure that the sensors’
output signals are reliable, unbiased, and freeof noise and interference.

DC VOLTAGE
Instrumentation Level DC Voltage

Data acquisition systems usually contain integrated-circuit signal conditioners and multiplexers at their input terminals. These ICs typically can’t handle more than about ±10 Vdc
input, although some may tolerate as much as±25 Vdc. One way around the limitation is toplace a voltage divider on the system’s input
terminals that reduces the voltage to meet theIC’s specifications. Some systems are manuallyswitched to the proper range, while others
select the range automatically.
In addition, after properly connecting thesensors, the item under test, and the dataacquisition system, apply power first to thedata acquisition system. Next energize thesensors and finally, the item under test. Some
data acquisition systems can be damaged
when voltages are applied to their inputs
while unpowered.

High DC Voltage

Voltages greater than about 24 Vdc are considered high voltage in data acquisition systemterms. Built-in or external voltage dividers andspecial signal conditioners are needed forinfo@mccdaq.com

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24

CHAPTER 4 Electrical Measurements

dividing tens or hundreds of volts down to just
10 V or less to protect the instrument’s input
circuitry. One critical consideration for thesafety of both the equipment and operators is
to use high-voltage insulation in wires, terminals, connectors, leads, sensors, and othercomponents to prevent leakage and arcing to
low-potential terminals and nearby objects.

Low DC Voltage

Compared to high voltages, successfullymeasuring low-level dc signals depends upon
appropriate wiring techniques between thesignal source and the data acquisition device.
Input amplifier stages in signal conditioning
equipment cannot distinguish between themeasured signal and a noise voltage coupled
into lead wires. When measuring signals lessthan one volt, shielded or unshielded twisted
pairs provide the best protection against noisevoltage pickup. Best wiring practices call forshields to be grounded at one end only,
preferably at the signal source.

AC VOLTAGE
Low AC Voltage

General-purpose data acquisition systems often
measure low-level ac voltages on the secondaryof loaded current transformers, step-down
potential transformers, clamp-on current
probes, current-sensing resistors, or sensors lacking galvanic isolation. All these low-level ac
voltages must have a low common-mode voltage component with respect to earth ground or
float with minimal ac leakage. Also, connect thevoltage source to earth ground when necessary.

Outlet Level AC Voltage

In the U.S. and Canada, utility wiring supports anominal 110 Vac, 60 Hz, single-phase voltage,
supplied with three wires; hot, neutral, andground. Europe typically uses 220 Vac, 50 Hz, andJapan supplies 100 Vac; 50 Hz in Eastern Japanand 60 Hz in Western Japan. In many factories,

Measurement Computing(508) 946-5100

220/230 to 460/480 Vac, single and three-phasevoltage, Wye and Delta configurations are also
distributed to electrical and electronic control
equipment. All these voltages are lethal and leads
should be connected or disconnected from thedata acquisition system and the equipment under
test with the power disconnected.
When connecting these voltage sources to dataacquisition systems, problems often crop up in
ground loops, which produce noise. Voltage
drops appear between the neutral wire andground for many reasons, some of which cannot
be eliminated easily or even found. Accepted
practices for shielding, grounding, and isolation
should be followed rigorously. Refer to Chapter
10 for more detailed information concerning
ground loops and interference.

High Voltage

Similar to dc voltage levels, data acquisition
system inputs typically can’t handle more than
±10 Vac peak. Higher ac voltages are attenuated and scaled with a signal conditioning
stage before they are processed. Voltages up to
2,000 V, peak-to-peak, can be measured with afully differential attenuator, calibrated tomatch the buffered differential inputs of a dataacquisition system. Attenuators are typically
voltage divider networks connected from thehigh side input terminal to a common ground.
For systems with a true differential input,
however, neither the input’s low nor high side
can be connected directly to ground. Twomatched resistor dividers attenuate the incoming
signal, one on the high side and the other on the
low side. Only the low side of the attenuator isgrounded. For example, an accessory device isrequired to provide a 200:1 attenuation without
affecting the amplifier’s high input impedance.
In a typical system, the two matched buffer
amplifiers, A and B (Figure 4.01), on the front
end of the data acquisition system draw almost
identical input bias current. Matched voltage
dividers placed on the high and low-side inputs

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)Data Acquisition Handbook

High-Voltage Adapter
High-voltage adapter≈50 kΩ

High-voltage adapter

C

5 MΩ

≈50 kΩ

A

Channel 1≈50 kΩ

C

5 MΩ

10 MΩ

B
5 MΩ

Data acquisition system

10 MΩ

A

10 MΩ
≈50 kΩ

Differential Input Attenuation

Data acquisition system

10 MΩ

25

B
5 MΩ

Channel 1

Fig. 4.01. A typical data acquisition system usuallyrequires an attenuator at its input to accept voltages
that exceed 10 V.

Fig. 4.02. A differential amplifier requires a resistor
at both inputs when running single-ended, but only
one resistor in a voltage-divider attenuator.

provide symmetrical current limiting from either
input and let the quasi-differential input of thedata acquisition system measure a lower voltage
4.01 (6.11)
referenced to analog Figurecommon.
Because theoutputs from amplifiers A and B are identical,
they can be considered a common-mode voltagethat the differential amplifier C rejects.

the divider on the high side input, but the low
side input has no matching drop. The difference
appears at the input to differential amplifier C
and produces an offset at its output.

This scheme attenuates the high and low
signals, and lets both ends float hundreds of
volts above ground. For example, the highinput can be 110 Vac while the low end has a
50 Vdc offset. After attenuation, the peakvoltage of the high side signal is 0.77 V andthe low side signal is 0.25 V, which are wellwithin the readable range of the data acquisition system’s differential inputs.
When a single-ended measurement is attemptedby attenuating only the high signal and pulling
the low signal to ground, two problems candevelop. First, when the data acquisition system
ground and the signal ground are different,tying them together forms a ground loop, which
can result in excessive current and destroy the
instrument or equipment. The second error
component comes from the bias currents Ib
flowing through the input pins of the two operational amplifiers, A and B shown in Figure 4.02.
The bias current produces a voltage drop throughMeasurement Computing(508) 946-5100

Effective, Peak, Average, and RMS
Voltage

Dc voltages and currents are relatively easy tomeasure and calculate with Ohm’s law. For
example, dc power is mathematically equivalent to the product of dc voltage and current
and generates a unique amount of heat energy
or performs a unique amount of work. By
comparison, power calculations for ac voltage
and current includes an additional factor, thecosine of the phase angle between them,called the power factor. When the voltage andcurrent waves are in phase, the cosine of theangle is unity, so the product of voltage andcurrent is a valid power value measured inwatts. When they are out of phase, however,the power factor is less than unity, so thepower generated is less than the simple productof voltage and current. In an extreme case, 90˚for example, the cosine of the phase angle iszero, so the wattage is also zero. However, thepower is then totally reactive, does not
generate heat (theoretically), and is simply
labeled VA (volt-amperes).

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26

CHAPTER 4 Electrical Measurements

Effective Voltage

V
141.4

(B) DC voltage

Amplitude (v)1000Effective voltage
Peak (B)(C)

AC sine wave
90˚

270˚

Peak (A)
450˚

t

(A)

-100
-141.4-V

Fig. 4.03. The numerical value for the peak voltageof ac must be 1.414 times that of dc to produce thesame heating effect.
EQN 4.01. DC Power

P=ExI
Where:
E and I are in phase
P = power, W
E = dc voltage, VI = dc current, A

heat. The value of ac voltage that produces anequivalent amount of work as the dc value iscalled the effective voltage and equals the dc
value multiplied by 1.414. In other words,
141.4 peak Vac produces the same amount ofheat in a given load, as does 100 Vdc.

RMS Voltage

In order to make ac and dc voltmeter scales
read the same value for the same amount ofwork done, the ac voltmeter scales are calibrated in rms (root mean square) voltage
values. When observing 100 Vac rms on themeter face, the voltage is actually 141.4 Vacpeak, and the amount of work it does equals
that of 100 Vdc (in the same load). Moreover,
for ac, the maximum (peak) positive excursion
and the maximum negative excursions are thesame; therefore the peak-to-peak value in thisexample is twice 141.4 or 282.8 Vp-p.
The rms voltage value of a sine wave is equal
to the effective value and is defined as thesquare root of the average of the squares of theinstantaneous voltages measured over a given
number of cycles.
EQN 4.03. Volts rms

EQN 4.02. AC PowerP = E x I x Cos φ
Where:
P = power, W
E = ac voltage, VI = ac current, A
φ = phase angle between voltage and current

Effective Voltage

V rms =

V 1 2 + V 2 2 + V 3 2 + ...V n 2

When the shape of the ac signal is not a nearperfect sine wave, ordinary vane-type voltmeters
will not indicate a true rms ac value. Special
meters and some data acquisition systems,
however, are designed to compensate for the
shape discrepancies and are called true rms
meters. They measure and indicate the true rms
value of the ac signal, regardless of the distortion. The average value of the ac voltage is zero,
because the positive half cycle magnitude equals
the negative half cycle of the wave shape.

The ac voltage wave shape follows a sine function (ideally), and because it continually rises
and falls over each cycle, the area under the
curve for ac is less than for dc overEquation
the same
4.03(peak) amplitude and time period. Thus,
CURRENT
100 Vac peak generates less heat than 100 Vdc
A Fundamental Problem
in the same load. (See Figure 4.03.) To compenVoltage drops are relatively easy to measure withsate for this, peak ac voltages must be higher
than dc to generate an equivalent amount of
most instruments because the difference in

Measurement Computing(508) 946-5100

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Data Acquisition Handbook

voltage appears between any two points in acircuit. The circuit does not need to be disturbed
or changed (provided the measuring instrument’s
impedance is high compared with the impedanceof the measured circuit) to make the measurement.
By comparison, simple direct-connection current
measurements are more troublesome, because
the current appears within a loop and the loop
must be opened to insert the ammeter. Moreover,
the ammeter contains a characteristic resistanceor impedance that most often changes the circuit
parameters and must be compensated to obtainaccurate measurements.

Typical Data Acquisition Input
+15
+
Vout

–
–15

13.8 V
commonmode

Some current measuring instruments, however,can couple inductively and have little or noaffect on the circuit parameters. Such probes
are designed for both ac and dc circuits.

Insertion Loss

Inserting an instrument or device in series withthe circuit causes the total series impedance tochange. In many applications, the added
Figure 4.04 Alternatecomponent or sensor requires some power tooperate or drops some voltage resulting in what
is called insertion loss. Therefore, some signal
power may be consumed in making the measurement, which must be compensated to provide amore accurate measurement.

Common-Mode Limitations

Most differential amplifiers used for signal conditioners in data acquisition systems can sustain
only a limited common mode voltage, typically
about ±10 Vdc. However, more expensive amplifiers or signal conditioners are available for
special applications that sustain common mode
voltages exceeding the 10 Vdc limit.
One application, for example involves measuring
the voltage drop across a calibrated shunt resistor
in an alternator charging circuit. When theamplifier input is connected to the shunt resistor
to measure mV scaled to charging current, thecommon mode voltage seen by the amplifier
input is the maximum battery voltage, nominally

Measurement Computing(508) 946-5100

27

+

50 A
50 mV
shunt

ALT
Alternator

Battery–

Fig. 4.04. The battery charging current in an automobile electrical system is typically measured with ahandheld diagnostic voltmeter across a shunt resistor,
which is totally isolated from the battery voltage.
However, when a data acquisition input amplifier isconnected to the shunt, the input also must sustain
the common mode voltage represented by the battery.

13.8 Vdc, but may be as high as 18.5 Vdc duringhigh-current charging. Therefore, the commonmode voltage rating for the data acquisition
system input must be greater than 18.5 Vdc.

ShuntsData acquisition instruments usually measurehigh-level dc currents (in the ampere range) as
a 50 or 100 mV voltage drop across a calibrated
shunt resistor. Figure 4.05 shows a special
application for differential measurements
where the three shunts have one end placed in
the common side of a three-motor circuit. This
arrangement provides a common ground thatcan be connected to an inexpensive monitoring
unit with a non-isolated, low-voltage input.
Unfortunately, not all circuits that need morethan one shunt will have one end of eachshunt conveniently connected to a common

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28

CHAPTER 4 Electrical Measurements

Ground Referenced Differential
Measurement
Channels
CH0 H
CH0 L
CH1 H

Shunt Resistor Current Measurement
Current
sourceiR

Supply

CH1 L

ADC

CH2 L

ShuntsAnalogCommonGood for a low
signal reference–+A
100 kΩ

CH2 H

Motor
Loads

Differential amplifier

Bad for a low
signal reference

Mux Board

Fig. 4.05. High dc currents
are often
measured
Figure4.06(3.06) withshunts using differential inputs sharing a common
ground.

point. In these instances, more expensive,
isolated analog input amplifiers arerecommended.
Data acquisition systems also measure lowlevel dc currents with shunts. Many sensors
output a standard 4 to 20 mA current, which
is linearly related to the quantity being
measured. The signal current passes throughthe shunt resistor, as shown in Figure 4.06,
and the voltage drop across the shunt provides
the input for the ADC.
Ohms law defines the nature of this simple
circuit where the product of resistance (R) andcurrent (I) yields the voltage: V = IR.
Consequently, when the data acquisition
system requires a 10 V input for a 20 mA fullscale reading, a 500 Ω resistor does the job:
EQN. 4.04. Shunt Resistor

R = V/I = 10 V/0.02 A = 500 Ω
The current loop provides higher noise immunity and accuracy than does a voltage source
when measuring with long leads of relatively

Measurement Computing(508) 946-5100

Fig. 4.06. Sensors with 4 to 20 mA outputs are
typically placed in series with a resistor to develop
a voltage. The resistor is then connected to the
input of a differential amplifier to measure thevoltage drop.

high resistance. In this case, the lead resistanceis part of the 500 Ω, so in a simple series circuit,the shunt resistor is often adjustable in order tocalibrate the loop to exactly 20 mA full scale.
The accuracy and stability of the measured
voltage across the shunt are only as accurate
and stable as the shunt resistor itself. Resistor
accuracies are commonly 5, 1, 0.5, 0.1 and
0.01% and come with a temperature coefficientthat specifies their stability over a specifictemperature range. In addition, a 0.1% resistor
usually has a lower temperature coefficient andbetter long-term stability than a resistor withlower accuracy because of its construction,
which is usually wire wound or metal film vs.
carbon film or carbon composition.

Current Transformers
High ac currents can be measured with shunts
under special circumstances, but most often
such direct connections to ac lines are
extremely dangerous to humans. One way to
get around the hazard is to use current transformers (CT) that isolate ac line voltages andreduce input current by a specified ratio.
(See Figure 4.07.) A 500:5 CT, for example,
has a 100:1 ratio and generates a secondary

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Data Acquisition Handbook

Current Transformers

C.T.
ratio
100:1

High-current
conductor

0.01 Ω

Vout = 50 mVrms

I = 500 Arms

Fig. 4.07. Current transformers are a convenientmeans to measure high ac currents. The voltage
dropped across the resistor is proportional to the current through the transformer.

current of 5 A with 500 A in the primary.
When a low value load resistor such as 0.01 Ω
connects to the output of the current transformer, the full load secondary current
produces 50 mVrms, which can be easily readwith the data acquisition system’s analog
input. Although this output voltage seems
low, higher resistance values will drive the CT
out of its calibrated range and reduce theaccuracy of the measurement. Moreover, a CT
rated at 2 VA can handle only a maximum
usable load resistance (including lead wires)of 0.08 Ω.
For safety, always make certain that the current
loop is closed before applying power. An opencircuit CT can generate many thousands of
volts at its terminals with lethal current levels.
For example, consider a 5,000:5 A current
transformer with a turns ration of 1,000 to 1,
and 10 Vac output signal developed across acalibrated low impedance load. The opencircuit secondary voltage is then 1,000 X 10 =10,000 Vac, a very dangerous level to beconnecting or disconnecting to a data acquisition system shunt resistor.

29

Hall-Effect Sensors

In 1879, Dr. Edwin Hall discovered the fundamental operating principle of the sensor
named after him. The basic Hall-effect sensor
operates under the influence of three parameters; current, voltage, and a magnetic field.
Hall passed a constant current through a gold
foil conductor while exposing the surface ofthe conductor to a magnetic field. A voltage
drop developed across the opposite sides of
the conductor with a magnitude dependent
on the property of the sensor material (Hallcoefficient) and proportional to the intensityof the magnetic field. (See Figure 4.08.)
A modern Hall device replaces the original
conductor with a thin sheet of semiconductormaterial. The output voltage appears between
two opposite edges of the sheet when the flat
surface is exposed to a magnetic field, and acalibrated current is injected across the othertwo opposite edges.
The Hall-effect sensor is ideal for measuring
magnetic fields or ac and dc power. For power
applications, it becomes a multiplying device.
It automatically multiplies the voltage andcurrent to yield power and includes the phase

Hall-Effect Sensor Flux Meter
Flux lines
Current
sourceSemiconductor

Voltmeter

Fig. 4.08. When either the voltage or current port isregulated and calibrated, the Hall-effect device canbe used as a magnetic flux sensor. The unregulated
port provides an output signal proportional to themagnitude of unknown flux density. However, aHall-effect proximity sensor or switch is a more
widely used component based on this principle.

Figure 4.08 (4.09 A)

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30

CHAPTER 4 Electrical Measurements

angle factor as an integral property. An electromagnetic coil generates the Beta field with its
core perpendicular to the surface of the thin
Hall element inside a shielded package. The
coil connects in series with the load to sense
load current, and the load voltage connects
across the semiconductor element as shown in
Figure 4.09. The resulting current vector drivesa readout device calibrated in watts or becomesa control signal to a power controller in aclosed-loop circuit. Because the Hall-element
material is temperature sensitive, a packaged
Hall-effect wattmeter contains temperaturecompensating circuits. The defining equation
for the output is the product of the vectors:
EQN: 4.05. Hall-Effect Sensor PowerP = E x I x Cos (φ)
Where:
P = power in the load, W
E = load voltage, V
I = load current, A
φ = phase angle between load voltage andload current

A Hall-effect sensor intended to measure amagnetic field is constructed such that the flat
surface of the sensor element is exposed to themagnetic field. A calibrated voltage or current isapplied to the edges of the semiconductor as
described above, and the unit is calibrated toaccurately measure a specific range of Beta
(magnetic flux) values. The same principle iswidely used in a Hall-effect switch, where themagnetic field bias may be applied with apermanent magnet and a slotted vanealternately interrupts the flux path. The outputof the sensor switches between a low voltage
and a high voltage as the flux path is interrupted.
Such a device had been used as a switch formany years in numerous automobile distributors
to replace the ignition points.

RESISTANCE
Basic Measurements

Ordinary ohmmeters measure resistance indirectly by injecting a small current into theresistor under test and measuring the resulting
voltage drop across its terminals. Precision

Hall-Effect Sensor Wattmeter
Load

Iron containing
magnetic flux
Hall device
IL = Load current

Voltmeter
Power calibrated inindicator Watts - PW

RC
VL = Load voltage

Line

Transformer

VW = Voltage
proportional to power

Flux
lines

PW = ILVL x Cos φ
where φ = phase angle between voltage and current

Fig. 4.09. A Hall-Effect sensor is an ideal wattmeter for measuring ac power. It inherently includes thephase angle between the voltage and current vectors as a multiplying factor to convert VA to W.

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Data Acquisition Handbook

reference resistors called multipliers scaleanalog meters to read in the middle portion of
its face, while digital ohmmeters scale either
manually or automatically to properly position
the decimal point. Digital multimeters (DMMs)
similarly apply a dc voltage across an unknown
resistor and a precision shunt resistor. The test
current generates a voltage drop across theshunt resistor, which is read by the ADC to
output a digital value of the unknown resistor.
Data acquisition systems, however, do not
typically read ohms directly; they measure
voltages and voltage ratios across known andunknown resistance dividers and computeequivalent resistor values.

Voltage Divider Method
Resistance voltage divider circuits can use either
single-ended or differential-input amplifiers.
(Refer to Figure 4.10.) The known reference
resistor should have a value that is close to theunknown value, and the source voltage should
be low enough to ensure that the current iswithin acceptable limits such as 1.0 mA or less.

Resistive Voltage Divider
ReferenceRk(known)

+

A

–

B
Out
Gnd

Ru
(unknown)

+

A

–

B

x1

Vs

Data
acquisition
systemVs
Source

x1

Vm

Ru
Vm
=
Rk
Vs – Vm

Fig. 4.10. Resistive voltage dividers work withsingle-ended or differential amplifiers and a reference resistor whose value is near the unknown
resistor’s value.

Measurement Computing(508) 946-5100

31

For example, when measuring an unknown
resistor near 1,000 Ω, select a 1,000 Ω reference
resistor and a supply of 2.0 Vdc. This produces
a current of 2/2,000 = 1 mA. A 1,000 Ω reference resistor keeps the current relatively low
and satisfactorily measures unknown values
down to the 100 Ω range. The unknown
resistor value is calculated from:
EQN. 4.06. Voltage Divider Resistor

Ru = (Rk)(Vm)/(Vs – Vm)
Where:
Vm = measured input voltage, V
Vs = source voltage channel, V
Rk = known resistance, Ω
Ru = unknown resistance, ΩFor example:
Where:
Vs = 2 V
Vm = 1.10 V
Rk = 1,000 Ω
Ru = unknown resistor, ΩThen:
Ru = (1 k)(1.10 V)/(2.0 – 1.10)
Ru = (1,100)/(0.90) = 1,222.2 ΩConstant Current/Voltage Drop

Current source/voltage drop methods tomeasure unknown resistances use both two
and four-wire circuits. Two-wire circuits should
be avoided for measuring very low to moderately low resistances because the lead wires
and connections can become part of the total
resistance (which is added to the unknown
value).

Small Deviations

Resistance measurement errors come from anumber of sources. The most common errors
in low resistance circuits for simple constantcurrent/voltage drop methods using two
wires come from the lead wire resistance andconnections added to the unknown resistance. Keeping the lead wires short and using

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32

CHAPTER 4 Electrical Measurements

four-wire circuits such as Kelvin connections
can minimize these errors. Errors in highresistance measurements can come from theshunt resistance of insulation, such as in
wiring and connector bodies. When using acurrent source and measuring the voltage
drop, use high input impedance measuring
devices, whether they are voltmeters or signal
conditioner inputs. For extremely highresistor measurements, use a voltage source
and measure the current using short leadwires and robust connections.

WHEATSTONE BRIDGE
Basic Principle

Wheatstone bridges often measure precise
resistances and extremely small resistancechanges. A typical bridge circuit is arranged ina symmetrical pattern of equal-value seriesparallel resistors across a power source asshown in Figure 4.11. When all resistors are
identical, the voltage drops across them areequal, and the voltage between nodes A and B
is zero. When one or more resistors are notequal, which is usually the resistor under test,
the bridge unbalances and a null meter placed
between the nodes measures the amount of
unbalance. An adjustable resistor or a potentiometer can be placed in one arm of the bridge
to manually rebalance the circuit. When thepotentiometer drives the meter back to zero,its value then equals that of the unknown
resistor. The potentiometer comes with a
precision, calibrated readout device that indicates its resistance, which is also the value ofthe unknown resistor.

Anderson Loop Alternative

The basic Anderson loop is often compared toa passive bridge circuit, but it provides higher

Anderson Loop

Wheatstone Bridge
Precision resistor
decade box

Current loop

Precision calibrated
dial or digital readout

Rw

1.010

A

Rx
unknown
resistor

B+
Excitation
–

Rw
Rw

Rw
Null
meter

R w1

Current
sourceR

Subtractor

R w2

Vg
V1

Gage

∆R

R w3

+
A3
–

R w4

Vout

Vout

VREF
RF

Fig. 4.11. An unknown resistor placed in one arm ofa Wheatstone bridge typically unbalances the circuit
symmetry and produces an offset voltage across nodes
A and B. But adjusting a precision, variable resistor in
the opposite arm brings the circuit back into balance
when it equals the unknown resistor value, which
Figure4.12
may be read on a calibrated
dial.

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V2

A2

Fig. 4.12. The Anderson loop uses a unique subtractor circuit, which eliminates errors arising from leadwire resistance. It measures only the sensing element’s resistance change.

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33

Data Acquisition Handbook

accuracy measurements with less excitation
voltage. Furthermore, it contains an amplifier,which puts it in the class of an active device.

The output responses of a multiple strain gagearrangement can be expressed as:
EQN 4.08. Multiple Strain Gage Output

The basic Anderson loop is a series circuitcomposed of one sensing element in series
with one reference element powered by aconstant-current source. A dual-differential
operational amplifier with high temperaturestability and high CMRR, called a subtractor,delivers the output response from the sensing
element. (Refer to Figure 4.12.) The advantage of this topology lies in its ability toignore changes in lead wire resistance and to
amplify only the output changes of thesensing element.

V1out – Vref = I∆R1
V2out – Vref = I∆R2
V3out – Vref = I∆R3
V4out – Vref = I∆R4
Where:
Rref = Rn, nominal gage resistance
Vnout = output voltage, V
I = constant current, A∆Rn = the change in resistance of the nthstrain gage, ΩVref = reference voltage, V

The output response Vout of a basic straingage circuit is defined by:Anderson Loop for Multiple strain Gages

EQN. 4.07. Anderson Loop Output

Current loop

Vout = A1V1 – A2V2
Vout = Ir (∆Rg)
Where:
A1 = amplification factor of stage 1
A2 = amplification factor of stage 2
V1 = voltage across sensing strain gage, V
V2 = voltage across reference strain gage, V
Ir = constant current, A∆Rg = strain gage resistance change, ΩThe circuit is not limited to only one sensor;
it can be expanded to contain numerous
sensors in series such as a strain rosette. It
may be connected to signal conditioners
with only six wires plus a common as illustrated in Figure 4.13. Additional advantages
include an output voltage that is linear foreach gage response, it requires only 1/4 thepower needed by the conventional bridge
circuit, and the loop output voltage is twicethat of a conventional strain gage. The factthat the output signal of the Anderson loop
is twice that of a strain gage Wheatstonebridge circuit accounts for a 6 dB increase insignal-to-noise ratio.

i,electricalcurrent
sourceAnderson
subtractors
+
Gage
V1
R1 + ∆R1 –

+

+
Gage
V2
R2 + ∆R2 –

+

+
Gage
V3
R3 + ∆R3 –

+

+
Gage
V4
R4 + ∆R4 –

+

V1out

–
V2out

–
V3out

–
V4out

–
Instrumentation
amplifiers

Rrefreference
impedance

+

Vref
–

Vref

Reference
input

AnalogcommonFig. 4.13. The Anderson loop may be connected tomultiple sensors in series such as in this strain
rosette.

Figure 4.13

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34

CHAPTER 4 Electrical Measurements

The significance of the output signal
voltage, I∆Rn, is that it is principally a function of the change in resistance or impedance of the sensing element, not it’s absoluteterminal resistance.

SINGLE-ENDED AND DIFFERENTIAL
MEASUREMENTS
Basic Principles

Data acquisition systems provide for both
single-ended and differential-input connections. The basic difference between the two is
the choice of the common connection foranalog voltage inputs. Single-ended multichannel measurements require that all voltages be referenced to the same, well-chosencommon node to prevent certain types ofmeasurement errors. Sometimes, however, anideal common point cannot be secured and adifferential input is needed.
Differential connections cancel or ignore
common-mode voltages and can measure thedifference between the two connected points.
The rejected common mode voltages can besteady dc levels or noise spikes. When givena choice, the differential input is preferred.

Differential Thermocouple Measurement
ThermocoupleA
ib

Differential
amplifierRb = 10 kΩ

Fig. 4.14. The resistor is necessary for at least one
input to provide the path to ground for amplifierbias currents.

When to Make Single-Ended
Measurements

When single-ended and differential input
measurements are offered in the same dataacquisition system, the differential inputs
usually consume two of the single-ended
inputs. That is, 16 single-ended input amplifier channels equal 8 differential-input amplifier channels. WhenFigure
noise 4.14
susceptibility andground loops are not a concern, single-ended
input measurements may be satisfactory.

When to Make Differential Measurements

Differential measurements call for a differential amplifier to measure the difference
voltage between the two input signal leads.
Differential amplifiers are preferred because

Measurement Computing

they provide greater noise rejection thansingle-ended amplifiers. Furthermore, some
sensors, particularly strain gages, requiremeasuring the voltage difference between thetwo signal leads. Figure 4.14 depicts a differential amplifier configured for a thermocouple. Although the amplifier measures thevoltage difference between the two inputs, apath to ground is required on at least one
input of the amplifiers. The 10 kΩ resistor
between the low side and ground provides apath for amplifier bias current. If the resistor
is absent or too high in value, the bias current
drives the input to one of the power supply
rail values, saturating the amplifier, which
exceeds its usual ±10 V common-mode range.

(508) 946-5100

The common-mode voltage generated by thebias current, ib, is:
EQN. 4.09. Common Mode Voltage

Vcm = 2ibRb
Where:
ib = bias current, A
Rb = bias resistor, ΩThe minimum value for Rb may be as low as ashort circuit (on one side only), which reduces toa single-ended input configuration. Otherwise,
under worse-case conditions, the voltage error in

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Data Acquisition Handbook

percent of the applied voltage due to source
resistance and Rb is:
EQN. 4.10. Voltage Error∆E(%) = 100

(

Rs
2R b

)

Differential Strain Gage Measurement
100 kΩ
R4

Where:∆E = error, %Rs = source resistance, ΩRb = bias resistor, Ω

R1
–
+A

Vexc
R3

This applies to Rb for a balanced circuit wherebias resistors are on both inputs. Values for Rb
typically range from 10 kΩ to 10 MΩ. If both
equations cannot be satisfied simultaneously,
the source resistance is too high to operate
within specifications.
Some amplifier inputs come with built-in bias
resistors for differential measurements, while
Figure 4.15
users supply the bias resistors
for extremely highimpedance inputs. However, an incorrect usersupplied bias resistor might dramatically lower
the input impedance in non-isolated systems
and load the circuits being measured. Other dataacquisition systems provide a suitable range ofresistors that can be switched to analog common(preventing users from selecting an incorrect
resistor). Alternatively, users can disconnect all
resistors when a dc path already exists at thesensor for bias current to flow from the amplifier
inputs to analog common.
Consider the strain gage in Figure 4.15. Whenthe excitation voltage is referenced to the samecommon as the differential amplifier, bias
resistors are not required because the strain
gage itself provides a current path to groundfor the instrumentation amplifier. However,when the instrumentation amplifier is floatingor referenced to a different common, at least
one bias resistor is required. A balanced termination with a 100 kΩ resistor can also be used.
Common-mode rejection ratio, CMRR, is a
critical amplifier specification and a measureof quality for instruments making differential

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35(508) 946-5100

R2+

100 kΩ–

Excitation

Fig. 4.15. When the strain gage reference voltage
and amplifier share the same common terminal,
switched bias resistors are not necessary.

measurements. A large CMRR is necessary toaccurately measure a small differential signal
on a large common mode voltage. The CMRR
specifies the maximum effect of the common
mode voltage on a differential measurement.
Figure 4.16 shows the output signal free ofthe common mode input noise after passing
through the differential amplifier. The twonoise spikes entering the amplifier are equalin phase and amplitude, so the difference
amplifier cancels them.

Differential Mode Noise Rejection
S1

+
Diff
–

S1 – S2

S2

Fig. 4.16. The differential amplifier rejects voltages
that are common to both lines.

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36

CHAPTER 4 Electrical Measurements

An amplifier’s CMRR is typically specified indecibels, dB, a ratio of two voltages that may
differ by a large magnitude, and determined by:
EQN. 4.11. Common Mode Rejection RatiodB = 20 log(V1/V2)
Where:
V1 = larger voltage, V
V2 = smaller voltage, V
For example, a ratio of 10 equals 20 dB, a ratioof 100 equals 40 dB and each additional factorof 10 adds another 20 dB to the sum.
Furthermore, an amplifier with a specificationof 100 dB corresponds to a ratio of 100,000, so
an instrument with a CMRR of 100 dB
measuring a signal on a common-mode voltageof 1 volt can have an error of 0.01 mV. If thecommon-mode voltage is larger, or if greater
accuracy is needed, an amplifier with a CMRR
of at least 120 dB would be needed, a ratio of
one million to one.
Calculate the CMRR for a differential amplifier
given in a data sheet:
Where Adiff = differential gain = 10
Acm = common mode gain = 0.001553
EQN. 4.12. Common Mode Rejection Ratio

CMRR = Adiff/Acm
CMRR = 10/0.001553
CMRR = 6,439
Find CMRR in dB:
EQN. 4.13. CMRR Example

CMRRdB = 20 log 6439 = 20(3.820) = 76.4 dB

High Common-Mode Voltage
Measurements

Often, a small voltage riding on a muchlarger voltage must be measured. For
example, when a thermocouple sits on one
terminal of a battery, the signal conditioner
input must be able to measure the millivolt

Measurement Computing(508) 946-5100

output of the thermocouple while
suppressing the battery voltage. When thecommon-mode voltage is less than 15 V, an
instrument amplifier with differential inputs
can read the thermocouple voltage while
ignoring the battery voltage. When thecommon-mode voltage is higher than 15 V,
however, an isolation amplifier is usuallyrequired. (Also refer to Figure 4.04.)
Although isolation does not protect amplifiers against excessive normal-mode input
voltage (voltage across a pair of inputs), it
does protect against excessive common-mode
voltage. Isolation eliminates a potentiallylarge and damaging current that would flowfrom the signal source due to the commonmode voltage.
Several isolation methods inherently include
high common-mode voltage rejection. Eachchannel can have an isolation amplifier, or agroup of channels (not isolated from eachother) can be multiplexed and digitized by an
ADC before the digital data are isolated.
Isolation barriers can be optical, magnetic, or
capacitive. The most common types are optocouplers composed of infrared (IR) light-emitting diodes (LEDs) detected by photodiodes on
the opposite side of a high-voltage, quartz
barrier. Optocouplers can transmit numerical
data in serial pulse trains or sine waves andpulse trains whose frequency or pulse width
can be modulated with analog signals. Varying
LED current, as shown in Fig. 4.17, can alsotransmit analog information. Magnetic barriers
using transformers and capacitive barriers are
generally internal and used in monolithic or
hybrid isolation amplifiers.
In frequency-coupled isolation, a highfrequency carrier signal is inductively or
capacitively coupled across the isolation
barrier. The signal is modulated on the input
side and demodulated on the output side to
reproduce the original input signal.

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7)

Data Acquisition Handbook

resistance does not introduce significant
errors. (See Figure 4.18.)

Optical Couplers
Vinput

R

–
+

–
+

Voutput
R

Photo
diode

LED
LED

Input
groundPhoto
diode
OutputgroundFig. 4.17. When making high common-mode voltage measurements, optical couplers provide a convenient and safe means for connecting the voltage todata acquisition systems.

Isolated ADCs and their accompanying signal
conditioning circuits float. The ADC converts
the input signal to a digital signal, and theinterface for transferring the digital code isdigitally isolated. (See Chapter 6.)

Examples of Kelvin-Type Measurements

An example of the Kelvin circuit illustrates anohmmeter composed of a current source and a
voltmeter. (See Figure 4.19.) The voltmetermeasures the voltage dropped by the load
resistor so the value of the resistor is simply V/I.
The accuracy of the calculated resistance in thisconfiguration is directly proportional to theaccuracy of the ammeter and voltmeter. The
resistance is calculated with Ohms law:
EQN: 4.14. Kelvin Connection Resistance

Ru = V/I
Where:
Ru = unknown resistance, ΩV = voltmeter reading, V
I = ammeter reading, A
Rw = lead resistanceVs = voltage source

Kelvin Connection

KELVIN CONNECTIONSRLBasic Principle
Wheatstone bridge circuits are accurate,
simple, sensitive, and convenient ohmmeters
to operate when the bridge, the unknown
resistance, and the readout are all located near
one another. But when the excitation andregulation circuit are located several meters
away from the bridge circuit, the leadresistance, RL, may cause an error in the
excitation voltage.
One way around the problem is to use a Kelvinconnection, a method that connects theexcitation and regulator circuits to the bridgewith four wires. Two wires carry the excitation
current, and the other two, the sense lines,
measure the excitation voltage at the bridgeand carry no current. Because the signal
conditioning circuit has high input impedance,
Figure 4.18
little current flows in the input leads, so their

Measurement Computing

37(508) 946-5100

RL

R1

Excitation andregulation++
Vexc Vsense
–
–

R4
RL

R2

R3
RL
RL
RL

+
–

A

Signalconditioning

Fig. 4.18. The Kelvin connection eliminates leadresistance from altering the excitation voltage byusing a second pair of wires. The measuring device
has relatively high input impedance so it does not
draw current and develop a voltage drop across its
leads that would add an error.

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38

CHAPTER 4 Electrical Measurements

Common Ohmmeter Circuits
Basic Ohmmeter Circuit
Rw

+
+

+

V
–

–

i

+
Ru
–

Is
Rm

Rw

–
V≈Ω

Although the load or unknown resistor may belocated a considerable distance from theammeter and voltmeter, the long voltmeter
leads in the Kelvin equivalent circuit do notdrop significant voltage because they draw
negligible current. Moreover, specially designed
Kelvin clips are available that connect to theload with minimal voltage drop at the jaw
hinges and contact surfaces.

Kelvin Connection: Ohmmeter Circuit

Vs

A

+

Rw

+

I
Ru

–

Rw

V
–

V≈Ω

Kelvin Connection for Remote Voltmeter
I

Vs

Rw

A

Rw

+
V

–

i≈0
Rw

Ru

I

Rw
V≈Ω

Voltmeter calibrated in ohms
Rm = Multiplier resistors
Rw = Lead resistance
Ru = Unknown resistor
Vs = Voltage source
Is = Current source­ ig. 4.19. The basic ohmmeter circuit contains aFcurrent source and multiplier resistors so the voltmeter scale may be calibrated in Ω. The Kelvin connection uses more than one set of leads to eliminate thelead resistance as a variable in the equation for both
local and remote measurements.

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39

Chapter 5
FUNDAMENTAL SIGNAL CONDITIONINGAMPLIFICATION

Data Acquisition Block Diagram

Data Acquisition Front Ends

Data acquisition systems differ from single ordual-channel instruments in several ways.
They can measure and store data collectedfrom hundreds of channels simultaneously.
However, the majority of systems containfrom eight to 32 channels, typically in multiples of eight. By comparison, a simple voltmeter that can select a measurement among
several different ranges can be considered adata acquisition system, but the need tomanually change voltage ranges and a lack ofdata storage hobbles its usefulness.
Figure 5.01 illustrates a simple data acquisition
system consisting of a switching network(multiplexer) and an analog-to-digital converter(ADC), both of which Figure
were 5.01
discussed in
Chapters 2 and 3. The main subject of thischapter, the instrumentation amplifier (IA), isplaced between the multiplexer and ADC. The
individual circuit blocks each have uniquecapabilities and limitations, which together
define the system performance.
The ADC is the last in a series of stages
between the analog domain and the digitizedsignal path. In any sampled-data system, suchas a multiplexed data acquisition system, asample-and-hold stage preceding the ADC isnecessary. The ADC cannot digitize a timevarying voltage to the full resolution of the
ADC unless the voltage changes relatively
slowly with respect to sample rate. Some ADCs
have internal sample-and-hold circuits or use

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AnaloginputsMux

IA
ADC
Digital data

Fig. 5.01. A simple data acquisition system is composed of a multiplexed input stage, followed by an
instrumentation amplifier that feeds one accurate
and relatively expensive ADC. This arrangement
saves the cost of multiple ADCs.

architectures that emulate the function of thesample-and-hold stage. The discussion that
follows assumes that the ADC block includes asuitable sample-and-hold circuit (either
internal or external to the chip) to stabilize theinput signal during the conversion period.
The primary parameters concerning ADCs indata acquisition systems are resolution andspeed. Data acquisition ADCs typically run
from 20 kHz to 1 MHz with resolutions of 16
to 24 bits, and have one of two types of inputs,
unipolar or bipolar. The unipolar-type typically ranges from 0 V to a positive or negative
voltage such as 5 V. The bipolar-type typicallyranges from a negative voltage to a positive
voltage of the same magnitude. Many dataacquisition systems can read bipolar or
unipolar voltages to the full resolution of the

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40

CHAPTER 5 Fundamental Signal Conditioning

ADC, which requires a level-shifting stage tolet bipolar signals use unipolar ADC inputs
and vice versa. For example, a typical 16-bit,
100 kHz ADC has an input range of -5 V to
+5 V and a full-scale count of 65,536. Zerovolts corresponds to a nominal 32,768 count.
If the number 65,536 divides the 10 V range,
the quotient is an LSB (least significant bit)magnitude of 153 µv.
Multiplexing through high source impedances
does not work well. The reason that low source
impedance is necessary in a multiplexed system is easily explained with a simple RC circuit shown in Figure 5.02. Multiplexers have asmall parasitic capacitance from all signal
inputs and outputs to analog common. These
small capacitance values affect measurement
accuracy when combined with source resistance and fast sampling rates. A simple RC
equivalent circuit consists of a dc voltage
source with a series resistance, a switch, and acapacitor. When the switch closes at T = 0, thevoltage source charges the capacitor throughthe resistance. When charging 100 pF through10 kΩ, the RC time constant is 1 µs. In a
10 µs time interval (of which 2 µs is available

Parasitic RC Time ConstantRsource

+

Ron

–

A

C

Input and Source Impedance

Rs

VADC =Ri
Vsig
Rs + Ri

Ri

Transducer

Fig. 5.03A. The sensor’s source impedance Rsshould be relatively small to increase the voltage
divider drop across Ri, the amplifier’s input. Thiscan substantially improve the signal-to-noise ratio
for mV range sensor signals.

Fig. 5.02. The source resistance should be as low as
possible to minimize the time constant of the MUX’s
Figure 5.03 A
parasitic capacitance C and series resistance R. An
excessively long time constant can adversely affectthe circuit’s measurement accuracy.

Measurement Computing

Figure 5.03A shows how system input impedance and the transducer’s source impedancecombine to form a voltage divider, which
reduces the voltage read by the ADC. The input
impedance of most input channels is 1 MΩ ormore, so it’s usually not a problem when thesource impedance is low. However, some transducers (piezoelectric, for example) have highsource impedance and should be used with a
special charge amplifier. In addition, multiplexing can greatly reduce a data acquisition
system’s effective input impedance. (See also
Chapter 3, Multiplexing, Input and Source
Impedance.) The charge injection effects areshown in Figure 5.03B.

Vsig

MUX

R

for settling time), the capacitor only charges to
86% of the value of the signal, which introduces a major error. But a 1 kΩ resistor lets thecapacitor easily charge to an accurate value in
20 time constants.

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Operational Amplifiers

Many sensors develop extremely low-level
output signals. The signals are usually too
small for applying directly to low-gain,

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Data Acquisition Handbook

41

which are configured easily to amplify or buffer
signals. Integrated operational amplifiers
contain many circuit components, but are
typically portrayed on schematic diagrams as asimple logical functional block. A few external
resistors and capacitors determine how theyfunction in the system. Their extreme versatility
makes them the universal analog building
block for signal conditioning.

MUX Charge Injection Effects
Drive signal

Mux output

Fig. 5.03B. Analog-switching devices can produce
spikes in the MUX output during level transitions in
the drive signal. This is called charge-injection effectand can be minimized with low source impedance.

multiplexed data acquisition system inputs, sosome amplification is necessary. Two common
examples of low-level sensors are thermocouples
and strain-gage bridges that typically deliver
full-scale outputs of less than 50 mV.
Most data acquisition systems use a number ofdifferent types of circuits to amplify the signal
before processing. Modern analog circuits
intended for these data acquisition systems
comprise basic integrated operational amplifiers,

Most operational amplifier stages are called
inverting or non-inverting. (See Figure 5.04.) A
simple equation relating to each configuration
provides the idealized circuit gains as a functionof the input and feedback resistors and capacitors.
Also, special cases of each configuration makeup the rest of the fundamental building blocks,
namely the unity-gain follower and thedifference amplifier.

Inverting Amplifier Stages

The inverter stage is the most basic operational
amplifier configuration. It simply accepts an
input signal referenced to common, amplifies
it, and inverts the polarity at the output
terminals. The open-loop gain of a typicaloperational amplifier is in the hundreds ofOperational Amplifiers
Rf

V in

+
–

V in

Ri

–
+

V out

A
Gain = –Rf
Ri

Inverting

V out

A

Ri

Rf
Gain =

Rf
Ri+1

Non-Inverting

Fig. 5.04. The two basic types of operational amplifiers are called inverting and non-inverting. The stage gain
equals the ratio between the feedback and input resistor values.

Figure 5.04
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42

CHAPTER 5 Fundamental Signal Conditioning

thousands. But the idealized amplifier used toderive the transfer function assumes a gain of
infinity to simplify its derivation without
introducing significant errors in calculating
the stage gain. With such a high stage gain, theinput voltage sees only the voltage divider
composed of Rf and Ri. The negative sign inthe transfer function indicates that the output
signal is the inverse polarity of the input.
Without deriving the transfer function, theoutput is calculated from:
EQN 5.01. Inverting Amplifier

Vo = –Vin(Rf/Ri)
Where:
Vo = output signal, V
Vin = input signal, V
Rf = feedback resistor, ΩRi = input resistor, ΩFor example, for a 500 mV input signal and adesired output of -5 V:
Vo = Vin(Rf/Ri)
Figure 5.05
Vo/Vin = Rf/Ri
5/0.50 = Rf/Ri = 10
Therefore, the ratio between input and feedback resistors should be 10, so Rf must be
100 kΩ when selecting a 10 kΩ resistor for Ri.
(See Figure 5.05.)

Inverting Amplifier StageIF = Ii
V(–) = V(+) = 0 V
virtual groundRi 10 kΩIi

Rf 100 kΩ

0A
Ed = 0 V+–

Vin = 0.5 V

–
+

Io

A

0A
RL

Vo = VRL

Fig. 5.05. The output polarity of the inverting amplifier is opposite to that of the input voltage. The closedloop amplification or gain of this stage is Acl = 10,
which is the ratio of Rf/Ri or 100 kΩ/10 kΩ.

The simplified transfer function is:
EQN 5.02. Non-Inverting Amplifier

Vo = Vin(Rf + Ri)/Ri
For the same 500 mV input signal,
Rf = 100 kΩ, and Ri = 10 kΩ:
Vo/Vin = (Rf + Ri)/Ri,
Vo = Vi(Rf + Ri)/Ri
Vo = 0.50(100 k + 10 k)/10 k
Vo = 0.50(110 k/10 k) = 0.50(11)Vo = 5.5 V

The maximum input signal that the amplifier
can handle without damage is usually about 2 V
less than the supply voltage. For example, when
the supply is ±15 Vdc, the input signal should
not exceed ±13 Vdc. This is the single mostcritical characteristic of the operational amplifier that limits its voltage handling ability. (See
Chapter 4, Electrical Measurements.)

The input voltage limitations discussed forinverting amplifiers applies equally well to the
non-inverting amplifier configuration.

Non-Inverting Amplifier Stages

Differential Amplifiers

The non-inverting amplifier is similar to theprevious circuit but the phase of the output
signal matches the input. Also, the gain equation simply depends on the voltage divider
composed of Rf and Ri. (See Figure 5.06.)

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Differential-input amplifiers offer someadvantages over inverting and non-inverting
amplifiers. It appears as a combination of theinverting and non-inverting amplifiers asshown in Figure 5.07. The input signal is

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Data Acquisition Handbook

I F = Ii
Rf 100 kΩ

Ri 10 kΩ

–
0V

Vin

+

Io

A

EQN 5.03. Differential Amplifier+IL

Vin = 0.5 V–

Vo = VRL + Vi

RL 10 kΩFig. 5.06. The input and output polarities of the
non-inverting amplifier are the same. The gain of thestage is Acl = 11 or (Rf + Ri)/Ri.

6

07

impressed between the operational amplifier’s
positive and negative input terminals andcan be isolated from common or a groundpin. The optional ground pin is the key tothe amplifier’s flexibility. The output signalof the differential input amplifier responds
only to the differential voltage that exists
between the two input terminals. The transfer
function for this amplifier is:

Non-Inverting Amplifier Stage

Ii

Differential AmplifierFeedback gain resistor
Rf
100 kΩ
100 kΩ
Ri
100 kΩ
+
V2
–

+
–

V1

Ri
Terminating
resistor

Optional
groundconnection

g=
–
+

Rf
Ri

A
RL

Rf
100 kΩVo =g(V1 - V2)

0.1% resistors

Fig. 5.07. The output voltage of the basic differential amplifier is the difference between the two
inputs, or Acl = g(V1-V2), where g is the gain factor.
Because all resistors in this example are of equalvalue, the gain is unity. However, a gain of 10 may
be obtained by making the feedback resistor 10 times
larger than the input resistors, under the conditions
that both feedback resistors are equal and the input
resistors are equal.

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43(508) 946-5100

Vo = (Rf/Ri)(V1 – V2)
For an input signal of 50 mV, Where:
V1 = 1.050 V and V2 = 1.000 V
Vo = (Rf/Ri)(V1 – V2)
Vo = (100 k/100 k)(0.05 V)
Vo = 0.05 V
For a gain of 10 where Rf = 100 k and Ri = 10 k:
Vo = (Rf/Ri)(V1 – V2)
Vo = (100 k/10 k)(0.05 V)
Vo = 0.50 V
The major benefit of the differential amplifier is
its ability to reject any voltages that are common to both inputs while amplifying the difference voltage. The voltages that are common toboth inputs are appropriately called common
mode voltages (Vcm or CMV). The commonmode voltage rejection quality can be demonstrated by connecting the two inputs together
and to a voltage source referenced to ground.
Although a voltage is present at both inputs,
the differential amplifier responds only to thedifference, which in this case is zero. The ideal
operational amplifier, then, yields zero output
volts under this arrangement. (See the following section on Instrumentation Amplifiers,
High Common-Mode Amplifiers.)

Programmable-Gain Amplifiers

Programmable gain amplifiers are typically
non-inverting operational amplifiers with adigitally controlled analog switch connected
to several resistors in its feedback loop. An

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44

CHAPTER 5 Fundamental Signal Conditioning

external computer or another logic or binary
signal controls the addressable inputs of theanalog switch so it selects a certain resistor
for particular gain. (See Figure 5.08.) The
data acquisition system’s signal conditioners
sense the input signal amplitude andautomatically send the proper binary code
to the programmable gain amplifier (PGA)
to increase the gain for a low signal, or
decrease the gain for a large signal. The
input signal then can be measured anddisplayed without distortion.

Programmable-Gain Amplifier

+
Vi

A

–
Iin(–) = 0 A+5 V

Three 1 kΩ
pull-upresistors
0
1
0

A2
A1
A0

TTL logic level,010, closes S2

S0
S1
S2
S3

4 kΩ G=1
2 kΩ G=2
1 kΩ G=4
1 kΩ G=8

R’F

R’i

S7

INSTRUMENTATION AMPLIFIERS
A Fundamental Problem

Because signal levels from some transducers
may be just a few microvolts, special problems
concerning ground loops and spurious interference frequently arise when amplifying
them. Other transducers provide output
signals from differential signal sources tominimize grounding problems and reduce theeffect of common-mode interfering signals.
Amplifiers used in these applications musthave extremely low input current, drift, andoffset voltage; stable and accurate voltage
gain; and high input impedance and commonmode rejection.
Although common integrated operational
amplifiers with several stages and extremely
tight resistor ratios are often used, speciallydesigned instrumentation amplifiers are
preferred for these applications. The highperformance operational amplifiers still use
basic circuits but ensure that they provide
extremely high common-mode rejection anddon’t need high precision matched resistors to
set the gain. Many instrumentation amplifiers
are designed for special applications andprovide unique features to increase theiraccuracy and stability for those applications.
For example, the functional block following
the switching network in a data acquisition
system, Figure 5.09, is an instrumentation
amplifier with several critical functions. It
rejects common-mode voltages, amplifies
signal voltages, minimizes the effect of themultiplexer on resistance, and drives the
ADC input.

High Common-Mode AmplifiersFig. 5.08. The non-inverting amplifier is configured
for programmable gain and controlled by the binary
input signals from a computer to the addressable
inputs of the analog switch.

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The common mode voltage is defined as thevoltage applied from analog common to both
inputs when the inputs are identical. (SeeFigure 5.10.) However, when the two input
voltages are different, 4.10 and 4.20 V for

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Data Acquisition Handbook

Instrument Amplifier–

VsignalVs+

IA

Vout = AvVsignal

Vcommon-mode

Fig. 5.09. An instrumentation amplifier is typicallya differential input operational amplifier with a highcommon-mode rejection ratio.

example, the common mode voltage, Vcm, is4.10 V and the differential voltage between thetwo is 0.10 V. Ideally, the instrumentation
amplifier ignores the common mode voltage
and amplifies only the difference between theHigh Common-Mode Amplifiers
Rf
100 kΩ
100 kΩ+
VCM–

Ri
100 kΩ
Ri

–
+

A
RL

VoCM ≈ 0

95.3 kΩRfCommon-mode
adjustment

0.1% resistors

Fig. 5.10. The common mode voltage rejection ismeasured with the two inputs shorted together anda voltage applied to the node. The potentiometer isthen adjusted for minimum (VoCM) output fromthe amplifier, signifying that the best balance wasreached between the two inputs.

Measurement Computing(508) 946-5100

45

two inputs. The degree to which the amplifier
rejects common mode voltages is given by aparameter called common mode rejection ratio
CMRR. (See Chapter 4, When to Make
Differential Measurements.) The ability of an
instrumentation amplifier (IA) to reject highcommon mode voltages is sometimes confusedwith its ability to reject high voltages. (See
Chapter 4, Electrical Measurements).
The signal voltages measured are frequently
much smaller than the maximum allowed
input of the system’s ADC. For example, a 0 to
100 mV signal is much smaller than the 0 to 5 V
range of a typical ADC. A gain of 50 is needed
to obtain the maximum practical resolution forthis measurement. Instrumentation amplifiers
are capable of gains from 1 to more than 10,000,
but in multiplexed systems, the gains are usually
restricted to a range of 1 to 1,000.
Measurement errors come from the non-ideal
ON resistance of analog switches added to theimpedance of any signal source. But theextremely high input impedance of the IA
minimizes this effect. The input stage of an IA
consists of two voltage followers, which havethe highest input impedance of any common
amplifier configuration. The high impedanceand extremely low bias current drawn from the
input signal generate a minimal voltage drop
across the analog switch sections and produce amore accurate signal for the IA input.
The instrumentation amplifier has low output
impedance, which is ideal for driving the ADC
input. The typical ADC does not have high orconstant input impedance, so the preceding
stage must provide a signal with the lowest
impedance practical.
Some instrumentation amplifiers have limitations concerning offset voltage, gain error,
limited bandwidth, and settling time. The
offset voltage and gain error can be calibrated
out as part of the measurement, but the bandwidth and settling time are parameters thatinfo@mccdaq.com

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46

CHAPTER 5 Fundamental Signal Conditioning

limit the frequencies of amplified signals andthe frequency at which the input switchingsystem can switch channels between signals. A
series of steady dc voltages applied to an instrumentation amplifier in rapid succession generates a difficult composite signal to amplify. The
settling time of the amplifier is the time necessary for the output to reach final amplitude towithin some small error (often 0.01%) after thesignal is applied to the input. In a system thatscans inputs at 100 kHz, the total time spentreading each channel is 10 µs. If analog-todigital conversion requires 8 µs, settling time of
the input signal to the required accuracy mustbe less than 2 µs.

Although calibrating a system can minimize
offset voltage and gain error, it is not always
necessary to do so. For example, an amplifier
with an offset voltage of 0.5 mV and a gain of
2 measuring a 2 V signal develops an error of
only 1 mV in 4 V on the output, or 0.025%.
By comparison, an offset of 0.5 mV and again of 50 measuring a 100 mV signal develop
an error of 25 mV in 5 V or 0.5%. Gain error
is similar. A stage gain error of 0.25% has agreater overall effect as gain increases
producing larger errors at higher gains andminimal errors at unity gain. System software
can generally handle known calibration
constants with mx+b routines, but some

Integrated Instrumentation Amplifiers
Input 1
Vi

+
–

+
0V

R in ~
– ∞Ω

–

A1
Ri1
R

+

R

Ri2
–
Input 2
Vi

+
–

0V

+

Sense

–
V’o

Rm

Rf1

A3

Rf2

Vo

Reference

A2

R in ~– ∞Ω
Buffered input
stage

Differential amplifier output
stage

Fig. 5.11. The instrumentation amplifier exhibits extremely high impedance to the inputs V1 and V2. Resistor
Rm adjusts the gain, and the single-ended output is a function of the difference between V1 and V2.

1
Measurement Computing(508) 946-5100

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47

Data Acquisition Handbook

measurements are not critical enough tojustify the effort.
Integrated instrumentation amplifiers are
high-quality op amps that contain internal
precision feedback networks. They are ideal
for measuring low-level signals in noisy
environments without error, and amplifying small signals in the midst of highcommon-mode voltages. Integrated instrumentation amplifiers are well suited for
direct connection to a wide variety of
sensors such as strain gages, thermocouples,
RTDs, current shunts, and load cells. They
are commonly configured with three op
amps – two differential inputs and one
differential output amplifier.
(See Figure
Figure 5.12_A_pt1
5.11.) Some have built-in gain settings of 1
to 100 and others are programmable.

1.5 H

1

0.5 H

V1

1.333 F

2

V2

1Ω
RL

1’

2’
0

Output (dB)Integrated Instrumentation Amplifiers

Butterworth Filter

-5

-2

-10
-15
-20

0.1

0.5

1.0

2.0

ωc (rps)

3.0

Fig. 5.12A. Butterworth filter and response
characteristics

Programmable-Gain Instrumentation
Amplifiers
A special class of instrumentation amplifiers,
called programmable-gain instrumentation
amplifiers (PGIA), switch between fixed gain
levels at high speeds Figure
for different
input sig5.12_A_pt2
nals delivered by the input switching system.
The same digital control circuitry that selects
the input channel also can select a gain
range. The principle of operation is the sameas that described above for Programmable
Gain Amplifiers.

Chebyshev Filter

V1

Measurement Computing(508) 946-5100

2

V2

1Ω
RL

0.912 F

2’ Wc (rps)1’

FILTERING

0
Gain (dB)

The three most common types of filters arecalled Butterworth, Chebyshev, and Bessel.
(See Figures 5.12A, B, and C.) Each type has
unique characteristics that make it more suitable for one application than another. All may
Figure 5.12_B_pt1
be used for high-pass, low-pass, band-pass,and band-reject applications, but they havedifferent response profiles. They may be usedin passive or active filter networks.

0.996 H

1

-10
-18

0.1

0.5

1

2

3

ωc (rps)

Fig. 5.12B. Chebyshev filter and response characteristics

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48

CHAPTER 5 Fundamental Signal Conditioning

Bessel Filter1
Rs

Odd order N

L2

1Ω

V1

C1

C2

CN

RL

Even order N
LN
1Ω

RL

1Ω

Gain (dB)

0
-10
-50
-100

-140

1k

10k

100k

ω(rps)

Fig. 5.12C. Bessel filter and response
characteristics

The Butterworth filter has a fairly flat responsein the pass-band for which it is intended anda steep attenuation rate. It works quite well fora step function, but shows a non-linear phase
response. Chebyshev filters have a steeper
attenuation than Butterworth, but develop
some ripple in the pass band and ring with astep response. The phase response is much
more non-linear than the Butterworth. Finally,
Bessel filters have the best step response andphase linearity. But to be most useful, Bessel
filters need to have a high order (number ofsections) to compensate for their slower rate ofattenuation beyond the cut-off frequency.

Low-Pass Filters

Low pass filters attenuate higher frequencies in
Figurevarying degrees dependingon5.13the number ofMeasurement Computing(508) 946-5100

stages and the magnitude of the high frequency
relative to the corner frequency. An amplifier
stage doesn’t need high bandwidth when themeasured signal is at a much lower frequency. In
fact, the design is intended to eliminate excessive
bandwidth in all circuits, which reduces noise.
One major benefit of individual signal conditioning stages for low-level sensors (as opposed tomultiplexed stages) is to include low-pass filtering
on a per-channel basis in the signal path. In a
multiplexed circuit (an amplifier which is being
shared by multiple low-level dc signals), the main
signal path generally cannot operate as a low-passfilter due to the fast settling time necessary inmultiplexed systems. (See Figure 5.13A.)
The best place for low-pass filters is in the individual signal path before buffering and multiplexing. (See Figure 5.13B.) For small signals,
amplifying with an instrumentation amplifierprior to filtering lets an active low-pass filter
operate at optimum signal-to-noise ratios.
Figure 5.13C illustrates a typical amplifier-filtermultiplexer configuration.

Filtered/Buffered Data
Acquisition System
IA

LPF

BufferMux

InputsIA

LPF

To ADC

Buffer

Fig.5.13A. An instrumentation amplifier, a lowpass filter, and a unity gain buffer amplifier placed
between the sensor output and MUX input
transforms the sensor’s high output impedance to a
low impedance. This is required by the MUX to
maintain accuracy.

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Data Acquisition Handbook

Simple RC Filter
Vin

R

+

C

–

A

To Mux49

the vibrating laminations of a power transformer
mounted to its frame. Moreover, a combinationof high-pass and low-pass filters may be used tocreate a notch filter to attenuate a narrow bandof frequencies, such as 50 to 60 Hz and theirfirst harmonic. A 3-pole high-pass filter isshown in Figure 5.14.

Fig.5.13B. Low-pass filters inserted in eachchannel as needed simultaneously reduce thebandwidth and noise while passing the targeted,
lower frequency signals.

High-Pass Filter
C1

C2

C3

High-Pass Filters

High-pass filters operate in reverse to low-passfilters. They attenuate the lower frequencies
and are needed when low-frequency interference
can mask high-frequency signals carrying thedesired information or data. Low frequency
electrical interference sometimes couples into
the system from 50 or 60 Hz power lines.
Similarly, when analyzing a machine forvibration, the desired signals can be corrupted
by low frequency mechanical interference from

Amplifier/Filter/MUX Combination
Figure 5.14
Ch 1

IA

Low-pass filter
per channel

Ch 2

IA

Low-pass filter
per channel

Ch 3

IA

Low-pass filter
per channel

Ch 4

IA

Low-pass filter
per channel

Switched bias resistors(per channel)

Mux

RI

R1

R2

R3 Vo

RL

Fig. 5.14. The high-pass filter is designed to havea lower corner frequency near zero and the cutoff
frequency at some higher value. The number ofcapacitor/resistor pairs determines the number ofpoles and the degree of cutoff sharpness.

Passive vs. Active Filters

To A/D

2

Channel address lines

Fig. 5.13C. A typical production type data acquisition system contains an instrumentation amplifier and a low-pass filter in each channel preceding
the MUX. The amplifier boosts low-level signals
sufficiently well to let the active filter work at an
optimum signal-to-noise ratio.

Measurement ComputingVi(508) 946-5100

Passive filters comprise discrete capacitors,
inductors, and resistors. As the frequencies
propagate through these networks, two problems arise: the desired signal is attenuated by arelatively small amount, and when connected
to a load, the original filtering characteristics
change. However, active filters circumventthese problems. (See Figure 5.15.) They
comprise operational amplifiers built withboth discrete and integrated resistors, capacitors, and inductors. They can provide theproper pass-band (or stop-band) capability
without loading the circuit, attenuating thedesired signals, or changing the original
filtering characteristics. The advantages ofsuch a filter is; it is easier to design, and it has
finer tuning, fewer parts count, and improved
filter characteristics.

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50

CHAPTER 5 Fundamental Signal Conditioning

Active Filters

R1
Vi

C

R2

C1
R1

–
+

A

Vo

Vi

C2

R2

–
+

Vo
ActualIdeal

Stopband

A

Vo
Ideal

PassbandFigure 5.15 E

fL

0

Vo

f

Passband

Stop

High-Pass Filter
C

0

fL

fr

Stop
ActualfH

f

R1

Band-Pass Filter

R2

Vi

–
+

A

Vo

Figure 5.15_F
Vo
Ideal
ActualPassband0

StopbandfH

f

Low-Pass Filter

Fig. 5.15. Passive filters tend to change in frequency-cutoff characteristics with a change in load. To
prevent this, an active device such as a transistor orop amp isolates the last pole from the load to maintain stable filter characteristics.

Measurement Computing(508) 946-5100

Switched-Capacitance Filters

Although active filters built around operational amplifiers are superior to passive filters,
they still contain both integrated and discrete
resistors. Integrated-circuit resistors occupy alarge space on the substrate and their values
can’t be made easily to high tolerances, either
in relative or absolute values. But capacitorswith virtually identical values can be formed
on integrated circuits more easily, and whenused in a switching mode, they can replace
the resistors in filters.
The switched-capacitor filter is a relatively
recent improvement over the traditional active
filter. James Clerk Maxwell compared a switchedcapacitor to a resistor in a treatise in 1892, but
only recently has the idea taken hold in a zerooffset electronic switch and a high input impedance amplifier. The switched-capacitor concept

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Data Acquisition Handbook

is now used in extremely complex and accurate
analog filter circuits.
The theory of operation is depicted in Figure
5.16. It shows that with S2 closed and S1
open, a charge from V2 accumulates on C.
Then, when S2 opens, S1 closes, and thecapacitor transfers the charge to V1. This
process repeats at a particular frequency, andthe charge becomes a current by definition,that is, current equals the transfer of charge
per unit time.

Switched-Capacitor Filters
V2

V1
S2

S1
C

Fig. 5.16. Because resistors have wider tolerancesand require more substrate area than capacitors, atechnique that uses multiple precision capacitors to
replace resistors in filters is called a switchedcapacitor circuit.

The derivation of the equation is beyond thescope of this handbook, but it can be shownthat the equivalent resistor may be determined by:capacitors. Moreover, the equivalent resistor’s
effective value is inversely proportional to thefrequency or the size of the capacitor.

ATTENUATION
Voltage Dividers
Most data acquisition system inputs canmeasure voltages only within a range of 5 to
10 V. Voltages higher than this must be attenuated. Straightforward resistive dividers caneasily attenuate any range of voltages (seeFigure 5.17), but two drawbacks complicate
this simple solution. First, voltage dividers
present substantially lower impedances to thesource than direct analog inputs. Second, their
output impedance is much too high for multiplexer inputs. For example, consider a 10:1
divider reading 50 V. If a 900 kΩ and a 100 kΩ
resistor are chosen to provide a 1 MΩ load tothe source, the impedance seen by the analog
multiplexer input is about 90 kΩ, still too highfor the multiplexed reading to be accurate.
When the values are both downsized by afactor of 100 so the output impedance is lessthan 1 kΩ, the input impedance seen by the

Attenuator/Buffer
Vin
R1

EQN 5.04 Switched-Capacitance Filters(V2 – V1)/i = 1/(fC) = R
Where:
V2 = voltage source 2, V
V1 = voltage source 1, Vi = equivalent current, A
f = clock frequency, Hz
C = capacitor, FR = equivalent resistor, Ω

900 kΩ

9 kΩ
Vout

ZS = Source
100 kΩ
R2

1 kΩ

com.

ZL = Mux input

RL

com.
1 Meg
90 kΩ

The above equation states that the switchedcapacitor is identical to a resistor within theconstraints of the clock frequency and fixed

51

ZS
ZL

10 kΩ
1 kΩ

Fig. 5.17. When the input signal exceeds about
10 V, the divider drops the excess voltage to prevent
input amplifier damage or saturation.

Figure 5.17
Measurement Computing(508) 946-5100

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52

CHAPTER 5 Fundamental Signal Conditioning

measured source is 10 kΩ, or 2 kΩ/V, whichmost instruments cannot tolerate in a voltage
measurement. Hence, simple attenuation is
not practical with multiplexed inputs.

Buffered Voltage Dividers
The low impedance loading effect of simple
voltage dividers can be overcome using
unity-gain buffer amplifiers on divider
outputs. A dedicated unity-gain buffer has
high-input impedance in the MΩ range anddoes not load down the source, as does the
network in the previous example. Moreover,the buffers’ output impedance is extremely

Buffered Voltage Dividers
Vin
RA
+
–

RB

Vout

A

RL
Vout =Vin

RB
V
RA + RB in

RA

+Vc

Vout
RB

RE

RL

Fig. 5.18. An op amp or a transistor serves as an
impedance matching buffer to prevent the load from
affecting the divider’s output voltage.

Measurement Computing(508) 946-5100

low, which is necessary for the multiplexed
analog input. (See Figure 5.18.)

Balanced Differential Dividers

Not all voltage divider networks connect to aground or a common reference point at one
end. Sometimes, a balanced differential divider
is a better solution for driving the data acquisition system’s input terminals. (See Figure
5.19.) In this case, the CMRR of the differential amplifier effectively reduces the commonmode noise that can develop between different
grounds in the system.

High-Voltage Dividers

Some data acquisition systems employ special
input modules containing high-voltage
dividers that can easily measure up to 1,200 V.
They are properly insulated to handle the highvoltage and have resistor networks to select anumber of different divider ratios. They also
contain internal trim potentiometers to calibrate the setup to extremely close tolerances.

Compensated Voltage Dividers andProbesVoltage divider ratios applied to dc voltages are
consistently accurate over relatively long
distances between the divider network and dataacquisition system input when the measurement technique eliminates the dc resistance of
the wiring and cables. These techniques include
a second set of input measuring leads separate
from those that apply power to the divider suchas the Kelvin connection discussed in Chapter 4.
Voltage dividers used on ac voltages, however,must always compensate for the effective capacitance between the conductors and ground orcommon, even when the frequency is as low as
60 Hz. It’s not uncommon to find that when theac voltages are calibrated to within 0.01% at thedivider network, the voltages reaching the dataacquisition system input terminals may be outof tolerance by as much as 5%. This is because

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Data Acquisition Handbook

53

Compensated High-Voltage DividersHi

2.49 M

10 V
2M
249 K

Compensation
capacitors

220 pF

249 K

220 pF

249 K
249 K

Lo

2.49 M

2M

50 V
100 V

Select divider ratios

+
A
–

+
A
–

100 V
50 V
10 V

Out

+
–A

Input channel (typ. of 8)

Fig. 5.19. A typical high-voltage input card for a data acquisition system contains a balanced differential input
and jumpers for selecting an input voltage range of 10 V, 50 V, or 100 V. The input circuit also compensates forlong lead wire capacitance that tends to form an ac voltage divider, which would reduce accuracy.

Figurethe 5.19
lead capacitance enters into the divider

equation. One way around the problem is toshunt the data acquisition input terminals (orthe divider network) with a compensating
capacitor. For example, oscilloscope probes
contain a variable capacitor precisely for thispurpose. The probe capacitor is adjusted tomatch the oscilloscope’s input impedance andas a result, it faithfully passes the leading edge ofthe oscilloscope’s built-in 1,000 Hz square-wave
generator without undershoot or overshoot.

voltages, break ground loops, and eliminate
source ground connections without subjecting
operators and equipment to the high voltage.
They also provide a safe interface in a hospital
between a patient and a monitor or between thesource and other electronic instruments andequipment. Yet other applications include precision bridge isolation amplifiers, photodiodeamplifiers, multiple-port thermocouple andsumming amplifiers, and isolated 4 to 20 mA
current-control loops.

ISOLATION

Isolation Amplifiers

When Isolation Is Required
Frequently, data acquisition system inputs mustmeasure low-level signals where relatively highvoltages are common, such as in motor controllers, transformers, and motor windings. In
these cases, isolation amplifiers can measure
low-level signals among high common-mode

Measurement Computing(508) 946-5100

Isolation amplifiers are divided into input andoutput sections, galvanically isolated from eachother. Several techniques provide the isolation;
the most widely used include capacitive, inductive, and optical means. The isolation voltage
rating is usually 1,200 to 1,500 Vac, at 60 Hzwith a typical input signal range of ±10 V. They
normally have a high isolation mode rejection

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54

CHAPTER 5 Fundamental Signal Conditioning(IMR) ratio of around 140 dB. Because theprimary job of relatively low-cost amplifiers is
to provide isolation, many come with unity
gain. But more expensive units are available
with adjustable or programmable gains. (SeeFigure 5.20.)
One benefit of an isolation amplifier is that it
eliminates ground loops. The input section’s
signal-return, or common connection is isolated
from the output signal ground connection. Also,
two different power supplies are used, Vcc1 andVcc2, one for each section, which further helps
isolate the amplifiers. (See Figure 5.21.)

Galvanic Isolation AmplifierHigh
voltage

200 VDC

Load

X

Signal
Com 2
Ext
Osc

C3
–
+

R2

A1

To ADC
andcomputer

–
–
VCC1

–
VCC2
Groundisolation

Isolation barrier

Fig. 5.21. Galvanic isolation can use any one ofseveral techniques to isolate the input from the output circuitry. The goal is to allow the device to withstand a large common mode voltage (CMV) between
the input and output signal and power grounds.

OutputIsolation barrier
Ia

Vin

+

C2
C1L

R1

Differential V
CC2isolation+
amplifier
G=1

200 VDC

C1
C1H

VCC1
+

Shunt

Isolation

Input

Output50 mV+–

Analog Isolation Modules

Analog isolation amplifiers use all three types
of isolation between input and output sections:capacitive, optical, and magnetic. One type of
capacitively coupled amplifier modulates the
input signal and couples it Figure
across5.21
a capacitive
barrier with a value determined by the duty

Input

1pF

A
1pF

1pF

C2H

Ia

1pF

A
Sense
Ib

C2L

X

Sense

C5
Ib
A2

R3
R4

C4

R5

–
+

S/H
G=1

Sense

Vout

Signal
Com 2
S/H
G=6

Adjustable
gain stages+V2 Gnd -V2+V1 Gnd -V1
1

2

Fig. 5.20. A differential isolation amplifier’s front end can float as high as the value of the common mode voltage (CMV) rating without damage or diminished accuracy. The isolation barrier in some signal conditioners canwithstand from 1,500 to 2,200 Vdc.

Figure 5.20
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Data Acquisition Handbook

cycle. (See Figure 5.22.) The output section
demodulates the signal, restores it to the original analog input equivalent, and filters the
ripple component, which resulted from thedemodulation process. After the input andoutput sections of the integrated circuit are
fabricated, a laser trims both stages to precisely
match their performance characteristics. Thenthey are mounted on either end of the package
separated by the isolation capacitors. Although
the schematic diagram of the isolation amplifier looks quite simple, it can contain up to 250
or more integrated transistors.

55

Capacitive Isolation Amplifier
Input

Output+VC
Duty cycle
modulator

Vin
-VC
Com 1
Gnd 1
+VCC1

Duty cycle
demodulator

Sync

C2
T1
Oscillator
driverRectifiers
Filters

Ps Gnd-VCC1

Another isolation amplifier optically couples
the input section to the output section throughan LED transmitter and receiver pair as shownin Figure 5.23. An ADC converts the input
signal to a time-averaged bit stream and transmits it to the output section via the LED. The
output section converts the digital signal backto an analog voltage and filters
to remove
Figureit 5.22
the ripple voltage.

Sense

C1

Vout
Com 2
-VCC2
Sync
+VCC2
EnableGnd 2Isolation

Fig. 5.22. The isolation barrier in this amplifierprotects both the signal path and the power supply
from CMV breakdown. The signal couples througha capacitor and the power through an isolationtransformer.

Optical Isolation Amplifier
Input

OutputIsolation

IREF1

Rin
+
–

+In

Vin

I
-In

IREF2

Input

Output

I

I

–
+

Rf

–

A
Opticalcouplers

+

D1

D2

A

Vout

Vout = IinRf

LED

Common 1

Common 2

Fig. 5.23. This is a simplified diagram of a unity gain current amplifier using optical couplers between input and output
stages to achieve isolation. The output current passing through the feedback resistor (Rf) generates the output voltage.

Figure 5.23
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CHAPTER 5 Fundamental Signal Conditioning

Magnetically coupled isolation amplifiers comein two types. One contains hybrid toroid transformers in both the signal and power paths,
and the other contains one coil that transmits
the signal across a barrier to a giant magnetoresistor (GMR) bridge circuit. (See Figure 5.24A.)
In the transformer type, Figure 5.24B, the rectified output of a pulse generator (T1) supplies
power to the input and output stages (T3).
Another winding of the transformer (T2) operates a modulator and demodulator that carry
the signal across the barrier. It provides from
1,000 to 3,500 Vdc isolation among the amplifier’s three grounds, as well as an isolated
output signal equal to the input signal withtotal galvanic isolation between input andoutput terminals.
The second type, the giant magnetoresistor
amplifier uses the same basic technology as
does high-speed hard disc drives. The coilgenerates a magnetic field with strength
proportional to its input drive current signal,and the dielectric GMR amplifies and conditions it. Ground potential variations at the
input do not generate current so they are notdetected by the magnetoresistor structure. As aresult, the output signal equals the input signal
with complete galvanic isolation. These units
are relatively inexpensive, and can withstandfrom 1,000 to 3,500 Vdc. Full-power signal
frequency response is less than 2 kHz, but
small signal response is as much as 30 kHz.

Digital Method of Isolation

Digital isolation packages are similar in many
respects to analog amplifiers. They transmit
digital data across the isolation barrier at rates
up to 80 Mbaud, and some can be programmed
to transmit data in either direction, that is,through input to output or output to input
terminals. Data, in the form of complementary pulses, couple across the barrier throughhigh-voltage capacitors or air-core inductors.
Faraday shields usually surround the inductors
or capacitors to prevent false triggering fromexternal fields. The receiver restores the pulses
to the original standard logic levels. As withanalog amplifiers, the power supplies for eachsection are also galvanically isolated. (SeeFigure 5.25.)

Magnetic Isolation Amplifier
Input

OutputVc

VαSignaltransmitted
by magnetic
field
Vin

A

Iin

Planarcoil

H

Galvanic
isolationby thin-film
dielectricA

Isolated
output

GMR resistors

Isolation

Fig. 5.24A. Magnetic couplers transfer signals through a magnetic field across a thin film dielectric. In this case,a magneto-resistance bridge circuit sensitive to the field exhibits a large change in resistance when exposed to themagnetic field from a small coil sitting above it.

Figure 5.24 A
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Data Acquisition Handbook

57

Transformer-Type Isolation Amplifier
Input

OutputInput
Demodulator

T1

Pulse
generatorA

Modulator

T2

OutputDemodulator

Input
power

Rectifiers
and filters

T3

Rectifiers
and filters

+Vs
–Vs

R2
R1

–
+

Vin
R3

–
+

Vout

A

Outputpower

Isolation

Fig. 5.24B. The transformer-coupled isolation amplifier uses separate power supplies for the input and output
stages, which are isolated by virtue of the individual transformer windings. This provides the input and output
stages with decoupled ground or common returns. In addition, the modulator/demodulator transfers the measured
signal across the barrier via other transformer windings for complete galvanic isolation between input and output.

Isolated Sensors
FigureInherently
5.24 BIn addition to directly measuring voltage, current, and resistance, which require somedegree of isolation, certain sensors that measure other quantities are inherently isolated byvirtue of their construction or principle ofoperation. The most widely used sensors measure position, velocity, pressure, temperature,acceleration, and proximity. They also use anumber of different devices to measure thesequantities, including potentiometers, LVDTs(linear variable differential transformers), optical devices, Hall-effect devices, magnetic
devices, and semiconductors.
Hall-effect devices, for example, measuremagnetic fields, and are electrically insulated
from the magnetic source that they are designed
to measure. The insulation may be air or
another material such as plastic or ceramic, andthe arrangement essentially isolates them fromground loops and high voltages. Figures 5.26A
and B illustrate two applications where Halleffect devices measure speed. The first senses

the alternating magnetic field directly from therevolving wheel. In the second application, apermanent magnet sitting behind the Hall

Digital Method of Isolation
VDD1
Data
in

Input

OutputEncodeDecode

Gnd

VDD2
Data
out

Gnd
Air coretransformer

Isolation

Fig. 5.25. Yet another isolation method specificallyintended for digital circuits employs a high-speed
CMOS encoder and decoder at the input and output,
coupled with a monolithic air-core transformer.

Figure 5.25

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58

CHAPTER 5 Fundamental Signal Conditioning

effect device supplies the magnetic field. The
gear teeth passing by the unit disturb the field,
and the Hall device senses the resulting fluctuations. In addition, Figure 4.08. shows a third
example, where the Hall-effect device is used ina wattmeter circuit.

Isolated Sensors – Hall EffectHall-effectsensorGap

Gap

Current transformers and potential transformers for measuring ac voltage and current
are also inherently isolated between primaryand secondary windings. (See Figure 5.27.)
Transformer insulation between primary andRotation

Vsens

Magnetic field
S

Tone
wheel
Rotation

Isolated Sensors – Hall EffectN

Permanent
magnetIout

IH

Magnetic pole

Conditioned output

IL

N
3 mV

S

Targetwheel

Hall-effectsensor0

Amplifier,comparator

Input

N
S

A

Figure 5.26 B

OutputBridge circuit

Conditioned Output

pt 1
Current

10 mA
5 mA

Wheel position

Fig. 5.26A. This Hall-Effect sensor is switchedwith a series of alternating magnets in a target
wheel. Each pass between N and S magnets changes
the sensor’s state.

Measurement Computing

t

-3 mV

(508) 946-5100

Fig. 5.26B. This Hall-Effect sensor configuration
uses a bias magnet and a tone wheel that modulates the magnetic field intensity to produce an
output signal.

secondary can be made to withstand thousands of volts and have extremely low leakage
values. The turns ratio also is easy to select for
stepping down a high voltage to a lower standard voltage of 5 to 10 Vac.
Other sensors include magnetic pickups
composed of wire coils wound around apermanent magnetic core. A ferrous metal
passing over one end of the coil disturbs themagnetic flux and generates a voltage at the coilterminals. The sensor does not require a separate
power supply, and the output voltage is typicallysmall enough to require only ordinary signal
conditioners. (See Figure 5.28.)

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Data Acquisition Handbook

Isolated Sensor – Current Transformer
To data acquisition system
Rs

Piezoelectric materials and strain gages are
typically used for measuring acceleration, andare inherently isolated from the objects on
which they are mounted by virtue of theirprotective housings. High-voltage insulationand magnetic shielding may be added to theAmmeter

Isolated Sensor –Variable Reluctance/Magnetic

AC
To ammeter orshunt resistor, Rs

Antilock brake
controller

Donut-shaped
current transformer
220 VAC
power
input

Variablereluctancewheel speed
sensorAC inputfrom sensor

50 to 100A
load

Gear
wheel

DC
reference
AC inputfrom sensor

Ratio 1:1

1 turn59

I
I
Common or return

Ratio 2:1

2 turns

1.70V

15 mA

1.0V

5 mA
Digital output signal

Figure 5.28 A

I
ITargetWire coil

MagnetAnalog in

Common or return

Ratio 4:1

4 turns
I

DigitaloutputVR transducer

Figure5.28 BI

ure 5.27 B,C,D

Signalprocessing

Common or returnFig. 5.27. Because it does not need a ground connection, a current transformer is isolated from boththe input and output of the signal conditioner.

Fig. 5.28. Variable reluctance sensors comprise acoil of wire wound around a magnetic core. As aferrous metal passes near one pole it disturbs themagnetic field and induces a small voltage in thecoil. In this example, the voltage is amplified,shaped, and converted to a digital signal for indicating vehicle wheel speed.

Figure 5.28 C
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CHAPTER 5 Fundamental Signal Conditioning

mounting base if needed in some rare applications. (See Chapter 7, Strain Measurements.)
LVDTs contain a modulator and demodulator,
either internally or externally, require somesmall dc power, and provide a small ac or dc
signal to the data acquisition system. Often
they are scaled to output 0 to 5 V. LVDTs canmeasure both position and acceleration. (See
Chapter 9, Displacement and Position Sensing.)
Optical devices such as encoders are widelyused in linear and rotary position sensors. The
possible configurations are many, but basically,
their principle of operation is based on the

Thermocouple Output Voltage
E

Millivolts

60

K

40

T

TYPE
E
J
K
R
S
T

The transfer function for many electronic
devices, which relates the input to output,
contains a nonlinear factor. In most cases thisfactor is small enough to be ignored. However,in some applications it must be compensatedeither in hardware or software. Thermocouples,for example, have a nonlinear relationshipSeebeck Coefficient PlotR

500˚

Why Linearization Is Needed

J

20
0

LINEARIZATION

1000˚
1500˚
Temperature ˚C

100
S

2000˚

2500˚

METALS+–
Chromel vs. Constantan
Iron
vs. Constantan
Chromel vs. Alumel
Platinum vs. Platinum
13% Rhodium
Platinum vs. Platinum
10% Rhodium
Copper
vs. Constantan

Fig. 5.29. Although some thermocouples must be
both thermally and electrically connected to thespecimen under test, many may be purchased withinsulated junctions, which isolates them from making high-voltage and ground-loop connections to thesignal conditioner.

Seebeck Coefficient µV/˚C

80interruption of a light beam between an opticaltransmitter and receiver. A revolving opaque
disc with multiple apertures placed between thetransmitter and receiver alternately lets lightthrough to generate pulses. Usually, LEDs
generate the light and a photo diode on the
opposite side detects the resulting pulses, which
are then counted. The pulses can indicate position or velocity. (See Chapter 9, Displacement
and Position Sensing.)

80
T

60

E
J
Linear region

40

K

20

R

0
-500˚

S
0˚

500˚

1000˚

Temperature ˚C

1500˚

2000˚

Fig. 5.30. The slope of the Seebeck coefficient plotted against temperature clearly illustrates that thethermocouple is a non-linear device.

Figure 5.30
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Data Acquisition Handbook

from input temperature to output voltage,
severe enough to require compensation.
Figure 5.29 shows the output voltage forseveral types of thermocouples plotted against
temperature. Thermocouple output voltages
are based on the Seebeck effect (discussed
further in Chapter 6). When the slope of theSeebeck coefficient is plotted vs. temperature,
the output response is clearly nonlinear as
shown in Figure 5.30. A linear device, bycomparison, would plot a straight horizontal
line. Only the K type thermocouple most
readily approaches a straight line in the range
from about 0 to 1,000˚C.
The curve shows that one constant scale factor isnot sufficient over the entire temperature range
for a particular thermocouple type to maintainadequate accuracy. Higher accuracy comes fromreading the thermocouple voltage with a voltmeter and applying it to the thermocouple tablefrom the National Institute of Standards andTechnology as shown in Figure 5.31.
A computer-based data acquisition system, incontrast, automates the temperature conversion
process using the thermocouple voltage reading61

and an algorithm to solve a polynomial equation.
The equation describing this relationship is:
EQN 5.05. Temperature Polynomial

T = ao + a1x + a2x2 + a3x3 + ...anxn
Where:
T = temperature, ˚C
x = thermocouple voltage, V
a = polynomial coefficients unique to eachtype of thermocouple
n = maximum order of the polynomial
The accuracy increases proportionally to theorder of n. For example, when n = 9, a ±1˚Caccuracy may be realized. But because highorder polynomials take time to process, lower
orders may be used over limited temperature
ranges to increase the processing speed.

Software Linearization
The polynomials are put to work in the dataacquisition system’s computer to calculate thereal temperature for the thermocouple voltage.
Typically, the computer program handles anested polynomial to speed the process rather
than compute the exponents directly. Nested

Temperature Conversion Table in ˚C (IPTS 1968)
mV

.00

.01

.02

.03

.04

.05

.06

.07

.08

.09

.10

mV

0.00
0.10
0.20
0.30
0.40

0.00
1.70
3.40
5.09
6.78

0.17
1.87
3.57
5.26
6.95

0.34
2.04
3.74
5.43
7.12

0.51
2.21
3.91
5.60
7.29

0.68
2.38
4.08
5.77
7.46

0.85
2.55
4.25
5.94
7.62

1.02
2.72
4.42
6.11
7.79

1.19
2.89
4.58
6.27
7.96

1.36
3.06
4.75
6.44
8.13

1.53
3.23
4.92
6.61
8.30

1.70
3.40
5.09
6.78
8.47

0.00
0.10
0.20
0.30
0.40

0.50
0.60
0.70
0.80
0.90

8.47
10.15
11.82
13.49
15.16

8.63
10.31
11.99
13.66
15.33

8.80
10.48
12.16
13.83
15.49

8.97
10.65
12.32
13.99
15.66

9.14
10.82
12.49
14.16
15.83

9.31
10.98
12.66
14.33
15.99

9.47
11.15
12.83
14.49
16.16

9.64
11.32
12.99
14.66
16.33

9.81
11.49
13.16
14.83
16.49

9.98
11.65
13.33
14.99
16.66

10.15
11.82
13.49
15.16
16.83

0.50
0.60
0.70
0.80
0.90

1.00
1.10
1.20
1.30
1.40

16.83
18.48
20.14
21.79
23.44

16.99
18.65
20.13
21.96
23.60

17.16
18.82
20.47
22.12
23.77

17.32
18.98
20.64
22.29
23.93

17.49
19.15
20.80
22.45
24.10

17.66
19.31
20.97
22.62
24.26

17.82
19.48
21.13
22.78
24.42

17.99
19.64
21.30
22.94
24.59

18.15
19.81
21.46
23.11
24.75

18.32
19.97
21.63
23.27
24.92

18.48
20.14
21.79
23.44
25.08

1.00
1.10
1.20
1.30
1.40

Fig. 5.31. Because thermocouple outputs are non-linear, a table is an accurate method for converting a voltage
reading to temperature for a specific type of thermocouple.

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CHAPTER 5 Fundamental Signal Conditioning

NIST Polynomial Coefficients

a0
a1
a2
a3
a4
a5
a6
a7
a8
a9

TYPE E

TYPE J

Nickel-10% Chromium(+)
VersusConstantan (–)

Iron (+)
VersusConstantan (–)

TYPE K

TYPE R

TYPE STYPE T-100˚C to 1000˚C±0.5˚C
9th order

0˚C to 760˚C±0.1˚C
5th order

0˚C to 1370˚C±0.7˚C
8th order

0˚C to 1000˚C±0.5˚C
8th order

0˚C to 1750˚C±1˚C
9th order-160˚C to 400˚C±0.5˚C
7th order

0.104967248
17189.45282-282639.0850
12695339.5-448703084.6
1.10866E + 10
-1.76807E + 11
1.71842E + 12
-9.19278E +12
2.06132E + 13

-0.048868252
19873.14503-218614.5353
11569199.78-264917531.4
2018441314

0.226584602
24152.10900
67233.4248
2210340.682-860963914.9
4.83506E + 10
-1.18452E + 12
1.38690E + 13
-6.33708E + 13

0.263632917
179075.491-48840341.37
1.90002E + 10
-4.82704E + 12
7.62091E + 14
-7.20026E + 16
3.71496E + 18
-8.03104E + 19

0.927763167
169526.5150-31568363.94
8990730663-1.63565E + 12
1.88027E + 14
-1.37241E + 16
6.17501E + 17
-1.56105E + 19
1.69535E + 20

0.100860910
25727.94369-767345.8295
78025595.81-9247486589
6.97688E + 11
-2.66192E + 13
3.94078E + 14

Copper (+)
Nickel-10% Chromium (+) Platinum-13% Rhodium (+) Platinum-10% Rhodium
Versus Nickel-5% (–)
VersusVersusVersus(Aluminum Silicon)
Platinum (–)
Platinum (–)
Constantan (–)

Temperature Conversion Equation: T = a0 + a1x + a2x2 + ...+ anxn
Nested Polynomial Form: T = a0 + x(a1 + x(a2 + x(a3 + x(a4 + a5x)))) (5th order)

Fig. 5.32. The NIST polynomial table is a more accurate means of calculating a linearizing function for aparticular thermocouple than a single coefficient, even though it uses a polynomial equation.

Hardware Linearization
Hardware also may be designed to accommodate the nonlinearity of a thermocouple, but
the circuitry becomes complex and expensive
in order to reduce its susceptibility to errors
from outside influences such as electrical noise
and temperature variations within the circuits.
The compensating circuitry is nonlinear, andcontains breakpoints programmed withdiodes, resistors, and reference voltages, allsubject to errors that are avoided in software
compensation methods. However, several
modules are commercially available with

Measurement Computing(508) 946-5100

Figure 5.33

excellent, stable, built-in linearizing circuits.
The thermocouple voltage is extremely small
and most signal conditioners concentrate lesson compensation and more on amplifying thesignal while rejecting common-mode noise.
Alternative digital hardware methods use alook-up table to convert the thermocouple
voltage to a corresponding temperature.

Curve Divided into Sectors
Temp

Figure 5.32 (6.04
polynomials
are B)
the only practical way of
dealing with complicated equations. Without
such techniques it is difficult to handle large
state tables that have more than a few hundred
entries. (See Figure 5.32.) Also, high order polynomials can be computed faster when thethermocouple characteristic curve can bedivided into several sectors and each sector
approximated by a third order polynomial as
shown in Figure 5.33.

aVoltage

Ta = bx + cx2 + dx3

Fig. 5.33. Although a computer usually finds thesolution to the NIST polynomial, breaking the curve
into sections representing lower order polynomials
can accelerate the process.

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Data Acquisition Handbook

CIRCUIT PROTECTION

63

the high voltage input stage from the solid-state
circuitry. (See Figure 5.34.)

Hazards to Instrumentation Circuits
Many data acquisition systems contain solidstate multiplexing circuits to rapidly scanmultiple input channels. Their inputs are
typically limited to less than 30 V and may bedamaged when exposed to higher voltages.
Other solid-state devices in a measurement
system including input amplifiers and bias
sources also are limited to low voltages. However,these inputs can be protected with a programmableattenuator and isolation amplifiers that isolate

Another consideration often overlooked isconnecting active inputs to an unpowered dataacquisition system. Common safety practice
calls for all signals connected to the input of
the unpowered data acquisition system to bedisconnected or their power removed.
Frequently, de-energized data acquisition
system signal conditioners have substantiallylower input impedances than when energized,
and even low voltage input signals higher than

Isolation from High Voltage
Isolation
Ch 0
Isolation
Ch 1

Out
Mux Addr
Enb

Isolation
Ch 2
InputsIsolation
Ch 3
Isolation
Ch 4
Isolation
Ch 5
Isolation
Ch 6
Isolation
Ch 7

I/O
Gain
config.
µP–15 +15+5

DC/DC DC/DC
5 VDC
To DC/DC converters
in each channel
Isolation
amplifier

Input

Programmableattenuator2
andamplifier
Isolated supply+15–15

Wide range
regulator

9-20
VDC
supplyBypassLow
passfilter2-bit opto-iso

DC/DC

To
analogmux

From
on-boardµP
5 VDC

Fig. 5.34. The isolated voltage-input modules let data acquisition systems isolate several channels of analog
input, up to 500 V channel-to-channel and channel-to-system.

e 5.34
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CHAPTER 5 Fundamental Signal Conditioning

0.5 Vdc can damage the signal conditioners’
input circuits.

Overload Protection
Circuit A

D1
1 kΩ

–

Input
signal+D2

Circuit B

A

Output

-V

2 kΩ
G

Input
signalOverload Protection

+V

D

FET
S

–
+

A

OutputFig. 5.35. Certain CMOS multiplexers may be protected against overvoltage destruction with diodeand resistor networks that shut down any parasitic
transistors in the device, limit the current input toa safe level, and shunt input signals to groundwhen the power supplies are turned off (Circuit A).
JFET transistors also protect multiplexer inputs
when they are connected as diodes. The JFET
clamps at about 0.6 V, protecting the sensitive op
amp input from destruction (Circuit B).

Several methods are used to protect signal
conditioner inputs from damage when
exposed to transients of 10 to 100 V. Often, a
1,000 Ω, current-limiting resistor is in series
with the input when no voltage is applied tothe instrument amplifiers’ input. For transient
input voltages to 3,000 V and above, a series
resistor and a transient voltage suppressor are
installed across the input terminals. (SeeFigure 5.35.)

ESD Protection

Electrostatic discharge damage is a special
problem that can be avoided by handlingindividual circuit boards carefully andproviding proper shielding. ESD comes fromthe static charge accumulated on manydifferent kinds of materials, which finds a
return to ground or a mass that attracts theexcess electrons. The charge can create potential differences that can eventually arc overlarge distances. An arc containing only several
microjoules of energy can destroy or damage
a semiconductor device. Grounding alone is
not sufficient to control ESD build up, it only
ensures all conductors are at the same potential. Controlling humidity to about 40% andslightly ionizing the air are the most effective
methods of controlling static charge.
A discharge can travel one foot in one nanosecond and could rise to 5 A. A number of devices
simulate conditions for static discharge protection, including a gun that generates pulses at a
fixed voltage and rate. Component testing usually begins with relatively low voltages andgradually progresses to higher values.

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65

Chapter 6
TEMPERATURE MEASUREMENTTHERMOCOUPLE BASICS
The Gradient Nature of Thermocouples

Thermocouples (TCs) are probably the mostwidely used and least understood of all temperature measuring devices. When connected inpairs, TCs are simple and efficient sensors that
output an extremely small dc voltage proportional to the temperature difference between
the two junctions in a closed thermoelectric
circuit. (See Figure 6.01.) One junction isnormally held at a constant reference temperature while the opposite junction is immersedin the environment to be measured. The principle of operation depends on the uniquevalue of thermal emf measured between the
open ends of the leads and the junction of two
dissimilar metals held at a specific temperature. The principle is called the Seebeck Effect,
named after the discoverer. The amount ofvoltage present at the open ends of the sensor
and the range of temperatures the device canmeasure depend on the Seebeck coefficient,which in turn depends upon the chemicalcomposition of the materials comprising thethermocouple wire. The Seebeck voltage iscalculated from:
Equation 6.01. Seebeck Voltage∆eAB = α∆T
Where:
eAB = Seebeck voltage
T = temperature at the thermocouple junction
α = Seebeck coefficient∆ = a small change in voltage corresponding
to a small change in temperature
Thermocouple junctions alone do not generate
voltages. The voltage or potential difference thatdevelops at the output (open) end is a function

Type T Basic Thermocouple Circuit

Reference
temperature

Type T

Cu -NiConstantan T1
T1 Copper
0˚C (32˚F)
Open tomeasurevoltage

CopperCopperCopperDVM

T2

Cu -NiConstantan
CopperCopperMeasuredenvironmental
temperature

T2-270˚C to +400˚C(-454˚F to +752˚F)

Digital Voltmeter
output proportional to thedifference between T1 and T2

Fig. 6.01. A basic thermocouple measurement system requires two sensors, one for the environment under measurement and the other, a reference junction, normally held to 0˚C (32˚F). Type-T is one of the dozen or morecommon thermocouples frequently used in general-purpose temperature measuring applications. It is made ofcopper and constantan metals and typically operates from –270 to +400˚C or –454 to +752˚F.

Figure6.01 (6.01A)
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66

CHAPTER 6 Temperature Measurement

also copper) and outputs another emf that must
be compensated.

of both the temperature of the junction T1 andthe temperature of the open end T1’. T1’ must be
held at a constant temperature, such as 0˚C, toensure that the open end voltage changes inproportion to the temperature changes in T1. In
principle, a TC can be made from any two
dissimilar metals such as nickel and iron. In
practice, however, only a few TC types havebecome standard because their temperature
coefficients are highly repeatable, they arerugged, and they output relatively large voltages.
The most common thermocouple types arecalled J, K, T, and E, followed by N28, N14, S, R,
and B. (See the table in Figure 6.02.) In theory,the junction temperature can be inferred fromthe Seebeck voltage by consulting standard
tables. In practice, however, this voltage cannot
be used directly because the thermocouple wire
connection to the copper terminal at themeasurement device itself constitutes athermocouple junction (unless the TC lead is

Cold-Junction CompensationA
cold-reference-junction
thermocouple
immersed in an actual ice-water bath andconnected in series with the measuring
thermocouple is the classical method used tocompensate the emf at the instrument terminals.
(See Figure 6.03.) In this example, both copper
leads connect to the instrument’s input
terminals. An alternative method uses a single
thermocouple with the copper/constantan
connection immersed in the reference ice waterbath, also represented in Figure 6.03. The
constantan/copper thermocouple junction J2 in
the ice bath contributes a small emf that subtracts
from the emf from thermocouple J1, so thevoltage measured at the instrument or dataacquisition system input terminals corresponds
accurately to the NIST tables. Likewise, theCommon Thermocouple Types
Type

+

Metal

–

+

Standard
color code

B

Platinum6% Rhodium

Platinum30% Rhodium

E

Nickel10% Chromium Constantan Violet

J

Iron

K

Nickel10% Chromium–Ω/double foot20 AWG–

Seebeck
coefficient
S(µV/˚C) @ T(˚C)˚C standard
wire error

NBS specified
materials range*
(˚C)

0.2

6

600

4.4 to 8.6

0 to 1820**

Red

0.71

58.5

0

1.7 to 4.4-270 to 1000

Constantan White

Red

0.36

50.2

0

1.1 to 2.9-210 to 760

Nickel Yellow

Red

0.59

39.4

0

1.1 to 2.9-270 to 1372

N (AWG 14) Nicrosil

Nisil

–

–

39

600

–

0 to 1300

N (AWG 28) Nicrosil

Nisil

–

–

26.2

0

–

-270 to 400

R

Platinum13% Rhodium

Platinum

–

0.19

11.5

600

1.4 to 3.8-50 to 1768

S

Platinum10% Rhodium

Platinum

–

0.19

10.3

600

1.4 to 3.8-50 to 1768

0.30

38

0

0.8 to 2.9-270 to 400

–

19.5

600

–

0 to 2320

T

CopperConstantan Blue

W-Re

Tungsten5% Rhenium

Tungsten26% Rhenium

Red–* Material range is for 8 AWG wire; decreases with deceasing wire size** Type B double-valued below 42˚C – curve fit specified only above 130˚C

Fig. 6.02. NIST’s (National Institute of Standards and Technology) thermocouple emf tables publish the emf output
of a6.02
thermocouple
Figure(6.01 B) based on a corresponding reference thermocouple junction held at 0˚C.

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02 A)

Data Acquisition Handbook

Alternate Thermocouple Ice BathIsothermal blockT1 – T2
CopperCopperEnvironment
J1
to bemeasured T1
Constantan
Copper/Constantan
thermocouple
Type T

J4

Cu
Copper J

T4
Data
acquisition
input
T3

Type J Thermocouple
J3

Cu

T3

Data
Acquisition
System
Cu

Fe

Fe
Same

J4

T4 Fe

3

T3 = T4

Isothermal block

Figure 6.04 (6.02B)

copper wires connected to the copper terminals
on the instrument’s isothermal block do notneed compensation because they are all copperat the same temperature. The voltage reading
comes entirely from the constantan/ copper
thermocouple wire.
The above example is a special case, however,
because one lead of the type-T thermocouple iscopper. A constantan/iron thermocouple, on
the other hand, needs further consideration.
(See Figure 6.04.) Here, J2 in the ice bath is held
constant, and J1 measures the environment.
Although J3 and J4 are effectively thermocouple junctions, they are at the same temperature on the isothermal block, so they output
equal and opposite voltages and thus cancel.
The net voltage is then the thermocouple J1
output representing T1, calibrated to the NIST
standard table. If the I/O block were notisothermal, copper wire leads would be added
between the input terminal and the copper/
iron leads, and the copper/iron junctions (J3and J4) would be held in an ice bath as well, as
illustrated in Figure 6.05.

Measurement Computing(508) 946-5100

J1
T1

Iron/Constantan
thermocouple
Type J
J2

Reference
ice-water bathFig. 6.03. Whether J2 is a purchased thermocouple
or not, the junction formed by the constantan andcopper lead wires at J2 must be placed in the ice bathfor temperature compensation.

C
C

J2

T2

67

T2
Cu

Fe
– + V3 = V4
V3

Reference
ice-water bath

J3 = J4 = Fe/Cu
T3 = T4
V3 – V4 = 0

Fig. 6.04. One lead of a type-T thermocouple iscopper, so it does not require temperature compensation
when connecting to copper terminals. A Type J
constantan/iron thermocouple, on the other hand,needs a closer look. J2 remains constant in the icebath, and J1 measures the environment. Although J3
and J4 are effectively thermocouple junctions, they areat the same temperature on the isothermal block, sothey generate equal and opposite voltages and cancel.
Without an isothermal block, copper wire leads would
be added between the input terminal and the copper/
iron leads, and the copper/iron junctions (J3 and J4)
would be held in an ice bath as well.

Software Compensation

Ice baths and multiple reference junctions inlarge test fixtures are nuisances to set up andmaintain, and fortunately they all can beeliminated. The ice bath can be ignored when
the temperature of the lead wires and thereference junction points (isothermal terminal
block at the instrument) are the same. The emf
correction needed at the terminals can bereferenced and compensated to the NIST standards through computer software.
When the ice baths are eliminated, cold junction
compensation (CJC) is still necessary, however,

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C)

68

CHAPTER 6 Temperature Measurement

Chromel/Alumel Thermocouple
Copper+CopperV–
CopperCopperIsothermal block

Chromel
+ T1
V1
– J1
Alumel

Cu

T2

J2 Fe

RH

T1

J3 Cu
Type J

J3
J2

Data Acquisition
SystemEliminating the Ice BathThermocouple
input
card

Reference
ice-water bathIntegrated temperature sensor

Fig. 6.05. The chromel and alumel wire connections at
the copper leads constitute additional thermocouple
junctions that must be held at the same, constanttemperature. They generate equal and opposite
potentials which prevent them from contributing to thevoltage output from the chromel/alumel thermocouple.

Fig. 6.06. A thermistor sensor placed near the leadwire connections is an alternative method of replacing
the ice bath. The measured temperature is thedifference between the thermocouple temperatureand the reference thermistor temperature.

Figure 6.06 (6.03A)
in order to obtain accurate thermocouple
measurements. The software has to read theisothermal block temperature. One techniquewidely used is a thermistor, mounted close to
the isothermal terminal block that connects tothe external thermocouple leads. No temperature
gradients are allowed in the region containing
the thermistor and terminals. (See Figure 6.06.)
The type of thermocouple employed ispreprogrammed for its respective channel, andthe dynamic input data for the software includes
the isothermal block temperature and themeasured environmental temperature. The
software uses the isothermal block temperatureand type of thermocouple to look up the valueof the measured temperature corresponding to
its voltage in a table, or it calculates thetemperature more quickly with a polynomial
equation. The method allows numerous
channels of thermocouples of various types tobe connected simultaneously while the computer
handles all the conversions automatically.

Measurement Computing(508) 946-5100

Hardware Compensation

Although a polynomial approach is faster thana look-up table, a hardware method is evenfaster, because the correct voltage is immediately available to be scanned. One method is
to insert a battery in the circuit to null theoffset voltage from the reference junction so
the net effect equals a 0˚C junction. A morepractical approach based on this principle isan “electronic ice point reference,” which
generates a compensating voltage as a function of the temperature sensing circuit poweredby a battery or similar voltage source. (SeeFigure 6.07A.) The voltage then corresponds toan equivalent reference junction at 0˚C.

Type Mixing
Thermocouple test systems often measure tens
to hundreds of points simultaneously. In order
to conveniently handle such large numbers ofchannels without the complication of separate,unique compensation TCs for each,

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(6.03B)

Data Acquisition Handbook

Hardware Ice-Bath Replacement
Cu

Fe
T1
J1

+
R1

C

Iron/Constantan
thermocouple
Type J

Cu

+

+

R2

SignalSupplyR c voltage

e
R3

RT

–

–

Data Acquisition Systeme varies as a function oftemperature sensitive resistor RT

Fig. 6.07A. A number of electronic circuits or
modules can replace the ice-bath. The temperaturesensitive resistor changes the calibrated value ofvoltage e in proportion to the amount of temperaturecompensation required.

69

thermocouple-scanning modules come withmultiple input channels and can accept any ofthe various types of thermocouples on any
channel, simultaneously. They contain special
copper-based input terminal blocks withnumerous cold junction compensation sensors
to ensure accurate readings, regardless of thesensor type used. Moreover, the module contains
a built-in automatic zeroing channel as well as
the cold-junction compensation channel.
Although measurement speed is relatively slower
than most other types of scanning modules, thereadings are accurate, low noise, stable, andcaptured in only ms. For example, one TC
channel can be measured in 3 ms, 14 TC
channels in 16 ms, and up to 56 channels in 61
ms. Typical measurement accuracies are better
than 0.7˚C, with channel-to-channel variation
typically less than 0.5˚C. (See Figure 6.07B.)

Linearization

After setting up the equivalent ice point reference emf in either hardware or software, theHardware Ice-Bath Replacement
Thermocouple Vin

56 Channels Total(Ch0 thru CH55)
High
CH0
Low (+)

CHs 1 thru 55
via mini-TC
connectors

Open TC
detection
–VB
+
IA
-

MUX
(-)

+V

+VB

Instrumentation
amplifier

Channel select lines–V

To A/D

Filterprecharge
+

+

CJC1
CJC2
CJC3
CJC4

MUX

Cold junction
sensors-

DEMUX

Fig. 6.07B. A typical input scanning module can accommodate up to 56 thermocouples of any type, and up to896 channels can be connected to one analog-to-digital mainframe.

Figure 6.07B

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70

CHAPTER 6 Temperature Measurement

measured thermocouple output must be converted to a temperature reading. Thermocouple
output is proportional to the temperature of the
TC junction, but is not perfectly linear over a
very wide range. (See Linearization, Chapter 5.)
The standard method for obtaining highconversion accuracy for any temperature uses
the value of the measured thermocouple voltage
plugged into a characteristic equation for thatparticular type thermocouple. The equation is apolynomial with an order of six to ten. The
NIST table in Figure 5.32 (page 62) lists thepolynomial coefficients for some common
thermocouples. The computer automaticallyhandles the calculation, but high-order polynomials take significant time to process. In orderto accelerate the calculation, the thermocouple
characteristic curve is divided into several
segments. Each segment is then approximatedby a lower order polynomial.
Analog circuits are employed occasionally tolinearize the curves, but when the polynomial
method is not used, the thermocouple output
frequently connects to the input of an ADC
where the correct voltage to temperature match
is obtained from a table stored in the computer’smemory. For example, one data acquisition
system TC card includes a software driver that
contains a temperature conversion library. It
Figure 6.08A (6.05 A)changes raw binary TC channels and CJC information into temperature readings. Some software packages for data acquisition systems supply
CJC information and automatically linearize thethermocouples connected to the system.

THERMOCOUPLE MEASUREMENTPITFALLS

in each channel can reduce the interfering acline noise. Common filters include resistors
and capacitors and active filters built around op
amps. Although a passive RC filter is inexpensive
and works well for analog circuits, it’s notrecommended for a multiplexed front end
because the multiplexer’s load can change thefilter’s characteristics. On the other hand, an
active filter composed of an op amp and a few
passive components works well, but it’s more
expensive and complex. Moreover, eachchannel must be calibrated to compensate forgain and offset errors. (See Figure 6.08.)

Passive Filters
R1
Input

C1

R2
C2

C3

OutputFig. 6.08A. Passive filters come in a variety of configurations to suit the application. They are built insingle or multiple sections to provide increasingly
steeper slopes for faster roll-off.

Active Filters
C

Input

R1

A

OutputR2

Noisy Environments

Because thermocouples generate a relatively
small voltage, noise is always an issue. The
most common source of noise is the utility
power lines (50 or 60 Hz). Because thermocouple
bandwidth is lower than 50 Hz, a simple filter

Fig. 6.08B. An active filter easily eliminates themost common sources of electrical noise that competes with the thermocouple signal such as the interference from 50/60 Hz supply lines.

Figure 6.08B (6.05 B)
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Data Acquisition Handbook

Additional Concerns
Thermocouple Assembly

Thermocouples are twisted pairs of dissimilar
wires that are soldered or welded together at thejunction. When not assembled properly, they
can produce a variety of errors. For example,
wires should not be twisted together to form ajunction; they should be soldered or welded.
However, solder is sufficient only at relatively
low temperatures, usually less than 200˚C. And
although soldering also introduces a third metal,
such as a lead/tin alloy, it will not likely introduce errors if both sides of the junction are at
the same temperature. Welding the junction ispreferred, but it must be done without changing
the wires’ characteristics. Commercially manufactured thermocouple junctions are typicallyjoined with capacitive discharge welders that
ensure uniformity and prevent contamination.
Thermocouples can become un-calibrated andindicate the wrong temperature when the physical makeup of the wire is altered. Then it cannot
meet the NIST standards. The change can comefrom a variety of sources, including exposure to(Open)RL
Leakage resistanceRs

Rs

To DVM

RL
T1

Rs

temperature extremes, cold working the metal,
stress placed on the cable when installed, vibration, or temperature gradients.
The output of the thermocouple also can change
when its insulation resistance decreases as thetemperature increases. The change is exponential and can produce a leakage resistance so lowthat it bypasses an open-thermocouple wire
detector circuit. In high-temperature applications using thin thermocouple wire, the insulation can degrade to the point of forming avirtual junction as illustrated in Figure 6.09. The
data acquisition system will then measure theoutput voltage of the virtual junction at T1
instead of the true junction at T2.
In addition, high temperatures can release
impurities and chemicals within thethermocouple wire insulation that diffuse into
the thermocouple metal and change its
characteristics. Then, the temperature vs.
voltage relationship deviates from the publishedvalues. Choose protective insulation intended
for high-temperature operation to minimize
these problems.

Thermocouple Isolation

Virtual Junction

To DVM

71

T2
Rs

Virtual junction

Fig. 6.09. A short circuit or an insulation failure
between the leads of a thermocouple can form an
unwanted, inadvertent thermocouple junction calleda virtual junction.

Thermocouple isolation reduces noise anderrors typically introduced by ground loops.
This is especially troublesome where numerous
thermocouples with long leads fasten directly
between an engine block (or another large
metal object) and the thermocouple-measurement instrument. They may reference different
grounds, and without isolation, the groundloop can introduce relatively large errors in
the readings.

Auto-Zero Correction

Subtracting the output of a shorted channel
from the measurement channel’s readings canminimize the effects of time and temperaturedrift on the system’s analog circuitry. Althoughextremely small, this drift can become a significant part of the low-level voltage supplied bya thermocouple.

6)
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5)

72

CHAPTER 6 Temperature Measurement

One effective method of subtracting theoffset due to drift is done in two steps. First,
the internal channel sequencer switches to areference node and stores the offset error
voltage on a capacitor. Next, as the thermocouple channel switches onto the analog
path, the stored error voltage is applied to theoffset correction input of a differential amplifier and automatically nulls out the offset.
(See Figure 6.10.)

couple opens or significantly changes resistance,
the capacitor charges and drives the input to oneof the voltage rails, which indicates a defective
thermocouple. (See Figure 6.11.)

Open Thermocouple Detector
One of 32 channels+0.13 V
10 MΩ

Auto-Zero CorrectionTC

Differential amplifierA

Ch x
Ch groundA

B

B

Amplifier's
offset correction

A

To A/DMUX switches
Auto zero Sampling
phase
phase
Control for "A" muxes
Control for "B" muxes

4700 pF
10 MΩ–0.13 VTo differential
amplifier

MUX
switches

Open TC detection circuit

Fig. 6.11. The thermocouple provides a short-circuit
path for dc around the capacitor, preventing it fromcharging through the resistors. When the thermocouple opens, due to rough handling or vibration, thecapacitor charges and drives the input amplifier tothe power supply rails, signaling a failure.

A/D sample

Figure 6.11 (6.08) (4.07)

Fig. 6.10. Auto-Zero Correction compensates foranalog circuitry drift over time and temperature.
Although small, the offset could approach themagnitude of the thermocouple signal.

Open Thermocouple Detection
Detecting open thermocouples easily andquickly is especially critical in systems withnumerous channels. Thermocouples tend to
break or increase resistance when exposed tovibration, poor handling, and long service time.
A simple open-thermocouple detection circuit
comprises a small capacitor placed across thethermocouple leads and driven with a low-level
current. The low impedance of the intact thermocouple presents a virtual short circuit to thecapacitor so it cannot charge. When a thermo-

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Galvanic Action

Some thermocouple insulating materials
contain dyes that form an electrolyte in thepresence of water. The electrolyte generates agalvanic voltage between the leads, which inturn, produces output signals hundreds of times
greater than the net open-circuit voltage. Thus,
good installation practice calls for shielding thethermocouple wires from high humidity and all
liquids to avoid such problems.

Thermal Shunting

An ideal thermocouple does not affect thetemperature of the device being measured, but
a real thermocouple has mass that when
added to the device under test can alter thetemperature measurement. Thermocouple
mass can be minimized with small diameter
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Data Acquisition Handbook

contamination, annealing, strain, and shunt
impedance. One solution to help ease thisproblem is to use the small thermocouple wireat the junction but add special, heavier
thermocouple extension wire to cover long
distances. The material used in these extension
wires has net open-circuit voltage coefficientssimilar to specific thermocouple types. Itsseries resistance is relatively low over long
distances, and it can be pulled through conduiteasier than premium grade thermocouple wire.
In addition to its practical size advantage,extension wire is less expensive than standard
thermocouple wire, especially platinum.
Despite these advantages, extension wire
generally operates over a much narrower
temperature range and is more likely to receive
mechanical stress. For these reasons, thetemperature gradient across the extension
wire should be kept to a minimum to ensure
accurate temperature measurements.

Improving Wire Calibration Accuracy

73

temperature coefficient, α, of 0.00385 Ω/Ω/˚C,
and chemically pure platinum has a coefficientof 0.00392 Ω/Ω/˚C.
The following equation shows the relationshipbetween the sensor’s relative change in resistance with a change in temperature at a specific
α and nominal sensor resistance.
Equation 6.02. RTD Temperature Coefficient∆R = αRo∆T
Where:
α = temperature coefficient, Ω/Ω/˚C
Ro = nominal sensor resistance at 0˚C, Ω∆T = change in temperature from 0˚C, C
A nominal 100 Ω platinum wire at 0˚C willchange resistance, either plus or minus, over aslope of 0.385 Ω/˚C. For example, a 10˚C risein temperature will change the output of thesensor from 100 Ω to 103.85 Ω, and a 10˚C fallin temperature will change the RTD resistanceto 96.15 Ω.

Basics of Resistance Temperature
Detectors

Because RTD sensor resistances and temperature coefficients are relatively small, lead wires
with a resistance as low as ten ohms and relatively high temperature coefficients canchange the data acquisition system’s calibration. The lead wire’s resistance change overtemperature can add to or subtract from the
RTD sensor’s output and produce appreciableerrors in temperature measurement.

Platinum RTD resistances range from about
10 Ω for a birdcage configuration to 10 kΩ fora film type, but the most common is 100 Ω at
0˚C. Commercial platinum wire has a standard

The resistance of the RTD (or any resistor) is
determined by passing a measured current
through it from a known voltage source. The
resistance is then calculated using Ohms Law.
To eliminate the measurement error contributed by lead wires, a second set of voltage
sensing leads should be connected to the sensor’s terminals and the opposite ends connected to corresponding sense terminals at thesignal conditioner. This is called a four-wire
RTD measurement. The sensor voltage is measured directly and eliminates the voltage dropin the current carrying leads.

Thermocouple wire is manufactured to NIST
specifications. Often, these specifications can be
met more accurately when the wire is calibrated
on site against a known temperature standard.

RTD MEASUREMENTS

RTDs are composed of metals with a high positive temperature coefficient of resistance. Most
RTDs are simply wire-wound or thin film resistors made of material with a known resistancevs. temperature relationship. Platinum is oneof the most widely used materials for RTDs.
They come in a wide range of accuracies, andthe most accurate are also used as NIST
temperature standards.

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74

CHAPTER 6 Temperature Measurement

Measurement Approaches
2, 3, and 4-Wire Configurations

Four-Wire RTD with Current Source

Five types of circuits are used for RTD measurements using two, three, and four lead wires:Two-wire with current source, four wire withcurrent source, three-wire with current source,four-wire with voltage source, and three-wirewith voltage source.

RL

Vm
Irtd

RL
RTD

Two-Wire RTD

High impedance
voltmeter or
DAQ input

Constant
current
RL

A

Vm

To
ADC

RL

Rline

RL = Line resistanceFig. 6.13. The four-wire RTD method with a current
supply eliminates the lead wire resistance as a sourceof error.
Ohmmeter

RTD

Figure 6.13 (4.12)
Rline

Fig. 6.12. The simplest arrangement for an RTD
measurement is a simple series circuit containing
only two wires connected to an ohmmeter.

Figure 6.12 shows a basic two-wire resistancemeasurement method. The RTD resistance ismeasured directly from the Ohmmeter. But
Figure 6.03
this connection is rarely used since the leadwire resistance and temperature coefficientmust be known. Often, both properties are notknown, nor are they convenient to measure
when setting up a test.
Figure 6.13 shows a basic four-wire measurement method using a current source. The RTD
resistance is V/A. This connection is moreaccurate than the two-wire method, but it
requires a high stability current source andfour lead wires. Because the high-impedance
voltmeter does not draw appreciable current,
the voltage across the RDT equals Vm.

Measurement Computing
Figure 6.14 (508) 946-5100

Equation 6.03. 4-Wire RTD With Current Source

Vm
I rtd

R rtd =

Where:
Rrtd = RTD resistance, Ω
Vm = Voltmeter reading, V
Irtd = RTD current, A
Figure 6.14 shows a three-wire measurement
technique using a current source. The symbols

Three-Wire RTD With Current Source
Irtd

RL

Va

S1

Constant
current

A

RTD
RL
RL

High impedance
voltmeter or
DAQ input

Vb

Vm

To
ADC

S2

MUX
switches
RL = Line resistanceFig. 6.14. The three-wire RTD method with a current supply is similar to the four-wire method. It
simply eliminates one additional wire. Measure Va
first, then measure Vb.

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Data Acquisition Handbook75

Va and Vb represent two voltages measured by
the high-impedance voltmeter in sequence
through switches (or a MUX), S1 and S2. The
RTD resistance is derived from Kirchhoff’s
voltage law and by solving two simultaneous
equations. (Illustrating the solution is beyond
the scope of this book.) The benefit of thisconnection over that shown in Figure 6.13 is
one less lead wire. However, this connection
assumes that the two current-carrying wires
have the same resistance.

Figure 6.16 shows a three-wire measurement
technique using a voltage source. The RTD
resistance is derived from Kirchhoff’s voltage
law and three simultaneous equations. The
voltage source can vary as long as the sense
resistor remains stable, and the circuit is accurate as long as the resistances of the twocurrent-carrying wires are the same.

Equation 6.04. 3-Wire RTD With Current SourceA

R rtd =

I rtd

Irtd

Four-Wire RTD With Voltage Source
Constant
voltagesource
RL

A

RL

RTD

RL
RL

RL = Line resistanceVaFigure6.16High impedance
S1
voltmeter or
DAQ input
Vb
Vm
S2
To
A
Vc
ADC
S3
Vd

Rs

V1000 Ω RTD
V500 Ω RTD
V100 Ω RTD

Precision
resistor

S4

MUX
switches

RL

Va

RTD
RL

RL

Vb

Not
used

Vc
Vd

Rs

RL = Line resistance

Precision
resistor

V500 Ω RTD

V100 Ω RTD

S1
S2
S3
S4

High impedance
voltmeter or
DAQ input
Vm
A
To
ADC

MUX
switches

Fig. 6.16. This is a variation of the four-wire circuitwith a voltage source and a stable sense resistor.
Equation 6.06. 3-Wire RTD With Voltage Source

R rtd =Rs (2Vb – Va – Vd)
Vd

The RTD output is more linear than the thermocouple, but its range is smaller. The
Callendar-Van Dusen equation is often used to
calculate the RTD resistance:
Equation 6.07. RTD Curve Fitting

Fig. 6.15. The four-wire RTD circuit with a voltage
source is more complex than the four-wire withcurrent source, but the voltage is allowed to vary
somewhat provided the sense resistor is stable.

Figure 6.06

Equation 6.05. 4-Wire RTD With Voltage Source

R rtd =

V1000 Ω RTD

Constant
voltagesource(Va – Vb)

Figure 6.15 shows a four-wire measurement
system using a voltage source. The RTD resistance also is derived from Kirchhoff’s voltage
law and four simultaneous equations based on
the four voltages, Va through Vd. The voltage
source in this circuit can vary somewhat aslong as the sense resistor remains stable.

Irtd

Three-Wire RTD With Voltage Source

Rs (Vb – Vc)
Vd[

RT=Ro+Ro α T– δ

T
T
T
T
–1 )(
–1
–β(
( 100
100 )
100 ) ( 100 )]3

Where:
RT = resistance at T, Ω
Ro = resistance at T = 0˚C, Ω
α = temperature coefficient at T = 0˚Cδ = 1.49 (for platinum)
β = 0, when T>0
β = 0.11 when T<0Equation 6.07

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76

CHAPTER 6 Temperature Measurement

A typical value for self-heating error is 1˚C/
mW in free air. An RTD immersed in a thermally conductive medium distributes this heatto the medium and the resulting error issmaller. The same RTD rises 0.1˚C/mW in air
flowing at one m/s. Using the minimum excitation current that provides the desired resolution, and using the largest physically practical
RTD will help reduce self-heating errors.

Type S Thermocouple vs. Platinum RTD

Because lower currents generate less heat,currents between 100 and 500 µA are typically used. This lowers the power dissipation
to 10 to 25 µW, which most applications
can tolerate. Further reducing the current
lowers accuracy because they become more
susceptible to noise and are more difficult tomeasure. But switching the current on only
when the measurement is made can reduce
the RTD’s heat to below 10 µW. In a multichannel system, for example, the excitation
current can be multiplexed, much like theanalog inputs. In a 16-channel system, the

12

.390
.344
.293

8
4
0

200

400
600
Temperature, °C

Resistance temperature
coefficient-RTD

An alternative method involves measuring
RTD resistances at four temperatures andsolving a 20th order polynomial equation
with these values. It provides more precise
data than does the α, δ, and β coefficients in
the Callendar-Van Dusen equation. The plotof the polynomial equation in Figure 6.17
shows the RTD to be more linear than thethermocouple when used below 800˚C, themaximum temperature for RTDs.

800

Equivalent linearities type S thermocouple
Platinum RTD

Fig. 6.17. Platinum is the material of choice for
RTDs and thermocouples because it is stable andresists corrosion. Here, the RTD is shown to be
more linear under temperatures of 800˚C than thethermocouple.

Self-Heating
Another source of error in RTD measurements
is resistive heating. The current, I, passing
through the RTD sensor, R, dissipates power,
P = I2R. For example, 1 mA through a 100 Ω
RTD generates 100 µW. This may seem insignificant, but it can raise the temperature ofsome RTDs a significant fraction of a degree. A
typical RTD can change 1˚C/mW by selfheating. When selecting smaller RTDs forfaster response times, consider that they also
can have larger self-heating errors.

Scanning Inputs

Constant-Current Scanning Module
Ch x

Constant current

A

RTD

MUX
switches

To A/DCh x +1

RTD
24 Channels

Fig. 6.18. The constant-current source is sequentially switched among the various RTD sensors tokeep them cooler over the measurement interval andprevent resistive-heating errors.

Figure 6.18 (6.15)

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Data Acquisition Handbook

77

current will only excite each RTD 1/16th of
the time, reducing the power delivered toeach RTD from 100% to only 6%.

noise immunity. But the disadvantage is thehigh cost of buying or building an extremely
stable constant current source.

Two practical methods for scanning an RTD
include constant current and ratiometric.
An example of a constant current circuit isshown in Figure 6.18. It’s an RTD scanning
module, which switches a single 500 µA
constant current source among 16 channels.
A series of front-end multiplexers direct thecurrent to each channel sequentially while
the measurement is being taken. Both threeand four wire connections are supported toaccommodate both types of RTDs. By
applying current to one RTD at a time,errors due to resistive heating become
negligible. Advantages of the constantcurrent method include simple circuits andBy contrast, the ratiometric method uses a
constant voltage source to provide a current,
Is, through the RTD and a resistor, Rd. Fourvoltage readings are taken for each RTD
channel, Va, Vb, Vc, and Vd. (See Figure 6.19.)

Ratiometric Four-Wire RTD
Constant
voltage
IS
Figure6.08 A

V1000 Ω RTD
V500 Ω RTD
V100 Ω RTD

Va
RL

Vb

RTD

Vc
RL
IS

High impedance
voltmeter or
DAQ input

A

Precision
resistor

MUX
switches

Fig. 6.19. Four voltage readings are taken foreach RTD channel. The precision resistor measures Is, the RTD current; Vb and Vc measure the
RTD voltage; and the RTD resistance equals
Figure 6.09
(Vb – Vc)/Is.

Measurement Computing

Equation 6.08. 4-Wire RTD Ratiometric Measurement

Vd
Rd
V rtd = V b – V c
V rtd
R rtd =IsIs =For a three-wire connection (Figure 6.20), thevoltage, Va – Vc, includes the voltage drop
across only one lead. Because the two extension wires to the transducer are made of thesame metal, assume that the drop in the firstwire is equal to the drop in the second wire.
Therefore, the voltage across the RTD and its
resistance is:
Equation 6.09. 3-Wire RTD Ratiometric Measurement

V rtd = V a – 2(V a – V b ) – V d
V rtd
R rtd = R d Vd()

Practical Precautions

Vd
Rd

The current, voltage, and resistance of the
RTD is:(508) 946-5100

RTDs require the same precautions that apply
to thermocouples, including using shields andtwisted-pair wire, proper sheathing, avoiding
stress and steep gradients, and using large
diameter extension wire. In addition, the RTD
is more fragile than the thermocouple andneeds to be protected during use. Also, thermal
shunting is a bigger concern for RTDs than forthermocouples because the mass of the RTD is
generally much larger. (See Figure 6.21.)

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7)

78

CHAPTER 6 Temperature Measurement

Ratiometric Three-Wire RTD
Constant
voltageA

IS

V1000 Ω RTD
V500 Ω RTD
V100 Ω RTD

Va
RL

Vb
Not
used

RTD

Vc

RL
IS

A

Vd
Rd

Precision
resistor

MUX
switcheschange per ˚C (several %) than RTDs, whichmakes them more sensitive.
Thermistors are generally composed of semiconductor materials or oxides of common
elements such as cobalt, copper, iron, manganese, magnesium, nickel, and others. They
typically come with 3 to 6 in. leads, encapsulated, and color-coded. They are available in arange of accuracies from ±15˚C to ±1˚C, with anominal resistance ranging from 2,000 to
10,000 Ω at 25˚C. A value of 2252 Ω iscommon and can be used with most instruments. A plot of the temperature vs. resistancecharacteristic curves is usually provided withthe device to determine the temperature froma known resistance. However, the devices arehighly non-linear and the following equation
may be used to calculate the temperature:
Equation 6.10. Thermistor Temperature

Fig. 6.20. The three-wire ratiometric circuit assumes
that both sense-wire resistances in the four-wire circuit are the same. The equation for calculating RTD
resistance simply accounts for it with a factor of two.

RTD Resistance Comparison:
Small Resistance vs. Large Resistance
Response time
Thermal shunting
Self-heating error

Small RTD
Fast
Low
High

Large RTD
Slow
Poor
Low

Fig. 6.21. Although smaller RTDs respond faster totemperature changes, they areFigure
more 6.10
susceptible toinaccuracy from self-heating.

THERMISTOR MEASUREMENTS
Basics of Thermistors

Thermistors are similar to RTDs in that theyalso change resistance between their terminals with a change in temperature. However,they can be made with either a positive or
negative temperature coefficient. In addition,
they have a much higher ratio of resistance

Measurement Computing(508) 946-5100

1 = A + B(log R) + C(log R) 3eeT
Where:
T = temperature, ˚K
A, B, and C = fitting constants
R = resistance, ΩThe constants A, B, and C are calculated fromthree simultaneous equations with known
data sets: Insert R1 and T1; R2 and T2; R3 andT3, then solve for A, B, and C. Interpolation
yields a solution accurate to ±0.01˚C or better.

Linearization

Some thermistor manufacturers supply devices
that provide a near-linear output. They use
multiple thermistors (positive and negative
coefficients) or a combination of thermistors
and metal film resistors in a single package.
When connected in certain networks, they
produce a linearly varying voltage or resistance proportional to temperature. A widelyused equation for the voltage divider shown in
Figure 6.22 is:

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Data Acquisition Handbook

Equation 6.11. Thermistor Voltage Divider

For the resistance mode, see Figure 6.23.

( R +R Ro )Equation 6.13. Thermistor Resistance ModeE out = E in

Where:
Eout is the voltage drop across R

Linearize Thermistor Output Voltage
Linear Voltage vs. Temperature

Linearizing
resistors
EIN

Ro EOUT(Positive slope)

R1

R2

79

Rt = MT + b
Where:
T = temperature in ˚C or ˚F
b = value of the total network resistance Rtin Ω when T = 0
M = slope, Ω per degree T in ˚C or ˚F,Ω/˚C or Ω/˚F

Linearize Thermistor Output ResistanceLinear Resistance vs. Temperature
R2

T1

T2

R EOUT(Negative slope)

R1

Temperature-measuring
thermistors

T1

Fig. 6.22. The compensating resistors in series
with the thermistors improve linearity near thecenter of the thermistor’s S-shaped characteristic
curve. This is where the sensitivity is the greatest,
and its operating temperatures can be extended tocover a wider range.

If R is a thermistor, and the output voltage isplotted against the temperature, the curve
resembles an S-shapewith6.23
a fairly
Figure(6.20) straightcenter portion. However, adding other resistors or thermistors to R linearizes the center
portion of the curve over a wider temperature
range. The linear section follows the equationof a straight line, Y = mX +b:
For the voltage mode:
Equation 6.12. Thermistor Voltage Mode

Eout = ±MT + b
Where:
T = temperature in ˚C or ˚F
b = value of Eout when T = 0
M = slope, volts per degree T in ˚C or ˚F,
V/˚C or V/˚F
Measurement Computing(508) 946-5100

R

RL
Load
resistance

T2

Temperaturesensing thermistors

Linearizing
resistors

Fig. 6.23. Compensating resistors in the network
linearize the resistance change vs. temperature in thesame manner as they do for the voltage mode.

Although a lot of research has gone into developing linear thermistors, most modern dataacquisition system controllers and software
handle the linearization, which makes hardware linearization methods virtually obsolete.

Stability

Thermistors are inherently and reasonably stabledevices, not normally subject to large changes innominal resistance with aging, nor with exposure to strong radiation fields. However,prolonged operation over 90˚C can change thetolerance of thermistors, particularly those withvalues less than 2,000 Ω. They are smaller andmore fragile than thermocouples and RTDs, sothey cannot tolerate much mishandling.

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80

CHAPTER 6 Temperature Measurement

Time Constant

The time required for a thermistor to reach 63%
of its final resistance value after being thrust into
a new temperature environment is called its time
constant. The time constant for an unprotected
thermistor placed in a liquid bath may range
from 1 to 2.5 sec. The same device exposed to an
air environment might require 10 sec, while aninsulated unit could require up to 25 sec. Seventime constants is a universally accepted value toconsider when the device has reached its plateau
or about 99% of its final value. Therefore, a device
in the liquid bath might take as long as 7 sec tostabilize, while the same device in air could take
125 seconds or more than two minutes.

Dissipation Factor

The power required to raise the temperature of athermistor 1˚C above the ambient is called thedissipation factor. It is typically in the mW range
for most devices. The maximum operating
temperature for a thermistor is about 150˚C.

Tolerance Curves

Manufacturers have not standardized on thermistor characteristic curves to the extent they
have for thermocouples and RTDs. Thermistors
are well suited to measuring temperature set
points, and each thermistor brand comes with

its unique curve which is often used to design
ON/OFF control circuits.

Measurement Approaches
Temperature Measurement

Wheatstone bridge: Thermistors provide accurate temperature measurements when used in
one leg of a Wheatstone bridge, even atconsiderable distances between the thermistor
and the bridge circuit. (See Figure 6.24A.) The
lead length is not a critical factor because thethermistor resistance is many times that of thelead wires. Numerous thermistors can be
widely distributed throughout the lab orfacility and switched into the data acquisition
system without significant voltage drops acrossthe switch contacts. (See Figure 6.24B.)
Differential thermometers: Two thermistors
can be used in a Wheatstone bridge to accurately measure the difference in temperature
between them. Thermistors can be attached toany heat conducting medium in a system atvarious points to measure the temperature
gradient along its length. Two or more thermistors may be placed in a room to measuretemperatures at several different elevations
using the same basic switching arrangement.

Thermistors in a Wheatstone Bridge

A. Wheatstone Bridge

B. Remote Thermistors

RT2

Thermistor
R1

RT

Vs
A
R3

SourceR2

Null meter or input todata acquisition system

RT1
room
temperatures

R1

selectorswitchRT3

R3

Null meter or input todata acquisition system

Vs
A
R2
SourceFig. 6.24. An accurate temperature sensor can be fashioned from a thermistor in one leg of a bridge circuit. Lead
length is not significant; so several sensors may be switched in and out of a single monitor without losing accuracy.
Two thermistors make a differential thermometer that can be used for measuring temperature changes along apiping system or between various elevations in a building to balance the heating and air conditioning unit.

Figure 6.24 B
Measurement Computing(508) 946-5100

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81

Chapter 7
STRAIN MEASUREMENTS

STRAIN BASICS

Strain gages are sensing devices used in avariety of physical test and measurement
applications. They change resistance at their
output terminals when stretched orcompressed. Because of this characteristic, thegages are typically bonded to the surface of asolid material and measure its minute dimensional changes when put in compression or
tension. Strain gages and strain gage principles
are often used in devices for measuring acceleration, pressure, tension, and force. Strain is adimensionless unit, defined as a change inlength per unit length. For example, if a 1 mlong bar stretches to 1.000002 m, the strain isdefined as 2 microstrains. Strain gages have acharacteristic gage factor, defined as thefractional change in resistance divided by thestrain. For example, 2 microstrain applied to agage with gage factor of 2 produces a fractional
resistance change of (2x2)10-6 = 4x10-6, or
4 µΩ. Common gage resistance values typicallyrange from 120 to 350 Ω, but some devices canbe as low as 30 Ω or as high as 3 kΩ.

STRAIN MEASUREMENTCONFIGURATIONS
Wheatstone Bridge

To make an accurate strain measurement,
extremely small resistance changes must be
measured. A Wheatstone bridge circuit is
widely used to convert the gage’s microstrain
into a voltage change that can be fed to the
input of the ADC. (See Figure 7.01.) When all

Full-Bridge Circuit

R1
+

R3
–

Vexc

VOUT

R1
R4

+

R2
R3

–
R2

MotionR4

Fig. 7.01. The full-bridge circuit provides the largest output with minimum errors. All four arms of the bridge are
active; two are in tension while the two on the opposite side are in compression.

Measurement Computing(508) 946-5100

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82

CHAPTER 7 Strain Measurements

four resistors in the bridge are absolutely
equal, the bridge is perfectly balanced andVout = 0. But when any one or more of theresistors change value by only a fractionalamount, the bridge produces a significant,measurable voltage. When used with an
instrument, a strain gage replaces one or moreof the resistors in the bridge, and as the strain
gage undergoes dimensional changes (because
it is bonded to a test specimen), it unbalances
the bridge and produces an output voltage
proportional to the strain.

Two gages are mounted on the surface under
tension and the other two are mounted on
the opposite surface under compression. As
the member deflects, the two gages in tension
increase in resistance while the other two
decrease, unbalancing the bridge andproducing an output proportional to thedisplacement. The bridge output voltage is
given by:
EQN: 7.01. Full-Bridge Output Voltage

Vo = (Vex)(X)
Where:
Vo = bridge output voltage, V
Vex = excitation voltage applied to thebridge, V
X = relative change in resistance, ∆R/R

Full-Bridge Circuits

Although half-bridge and quarter bridge
circuits are often used, the full bridge circuit is
the optimal configuration for strain gages. It
provides the highest sensitivity and the fewest
error components, and because the full bridge
produces the highest output, noise is a lesssignificant factor in the measurement. For
these reasons, the full bridge is recommended
when possible.

The bridge nulls out potential error factors
such as temperature changes because all four
strain gages have the same temperature coefficient and are located in close proximity on
the specimen. The resistance of the lead wire
does not affect the accuracy of the measurement as long as the input amplifier has highA full bridge contains four strain gages
mounted on a test member. (See Figure 7.01.)

Half-Bridge Circuit
Bridgecompletion resistors

Strain
gages

R1

R3

+

–

Vexc
–

VOUT

R1

+

MotionR2
R2

R4

Fig. 7.02. In a half-bridge circuit, only two arms are active. Two strain gages are on the specimen while the two
fixed resistors that complete the bridge are not.

igure 7.02 (5.02)

Measurement Computing(508) 946-5100

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Data Acquisition Handbook

input impedance. For example, an amplifier
with a 100 MΩ input impedance produces
negligible current flow through the measurement leads, minimizing voltage drops due tolead resistance.

83

coefficients among bridge completion resistors and strain gages are different and theresistances do not change proportionallywith temperature. Furthermore, bridge
completion resistors are not usually locatednear the strain gages, so temperature differences contribute additional errors. In systems
with long lead wires, the bridge completion
resistors should be attached close to thegages, but this may not always be practicaldue to test fixture limitations or otherphysical conditions.

Half-Bridge Circuits
When physical conditions do not allow
mounting a full-bridge gage, a half bridge might
fit. Typically, two strain gages are mounted on atest member, and two discrete resistors complete
the bridge. The output voltage is:
EQN. 7.02. Half-Bridge Output Voltage

Quarter-Bridge Circuits

Vo = Vex (X/2)
Where:
Vo = bridge output voltage, V
Vex = excitation voltage applied to thebridge, V
X = relative change in resistance, ∆R/R

A quarter-bridge circuit uses one strain gageand three bridge completion resistors. The
output voltage is:
EQN. 7.03. Quarter-Bridge Output Voltage

Vo = Vex (X/4)
Where:
Vo = bridge output voltage, V
Vex = excitation voltage applied to thebridge, V
X = relative change in resistance, ∆R/R

For a large ∆R, half bridge and quarter-bridge
circuits can introduce an additional nonlinearity error. (See Figure 7.02). Also, the readings are not accurate when the temperature

Quarter-Bridge Circuit
Straingages

Bridgecompletion resistors

R1

R3

+

–

Vexc
–

VOUT

R1

+

MotionR4
R2

Fig. 7.03. A quarter-bridge circuit uses only one active arm and is the least sensitive of the three types. It is also
the most prone to noise and errors.

Figure 7.03 (5.03)

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84

CHAPTER 7 Strain Measurements

This arrangement has the smallest output, so
noise is a potential problem. Furthermore, all
the error sources and limitations in the halfbridge apply to the quarter bridge circuit. (SeeFigure 7.03.)

Excitation Source
Accurate measurements depend on a stable,
regulated, and low-noise excitation source. A
regulated source is necessary because theoutput voltage of a strain gage is also proportional to the excitation voltage. Therefore,
fluctuations in the excitation voltage produce
inaccurate output voltages.
An ideal data acquisition system provides an
excitation source for each channel, independently adjustable from 0.5 to 10.5 V with acurrent limit of 100 mA. An excitation
voltage, V, used with a strain gage of resistance, R, requires a current of I = V/R. The
resistance of a Wheatstone bridge measured
between any two symmetrical terminals
equals the value of one of the resistance arms.
For example, four 350 Ω arms make a 350 Ω
bridge. The load current equals the excitation
voltage divided by the bridge resistance; inthis case, 10 V/350 = 0.029 A = 29 mA.

Heating
Resistive heating in strain gages also should beconsidered because the gages respond totemperature as well as stress. In most standard
Figure 7.04 (5.04)
circuits, the heat that each gage dissipates isless than 100 mW, so it’s usually not a
problem. This is especially true when thestrain gage is bonded to a material that
conducts heat quickly, such as metal. However,
because materials such as wood, plastic, or
glass do not conduct heat away as rapidly, use
the lowest excitation voltage possible without
introducing noise problems. Also, heat canbecome a problem when the strain gages areuncommonly small, or numerous gages occupy
a limited space.

Measurement Computing(508) 946-5100

Consider a Kelvin connection for applying theexcitation voltage. Because the excitation
leads carry a small current, they drop a correspondingly small voltage; V = I/RL, which
reduces the voltage reaching the bridge terminals. As illustrated in Figure 7.04, Kelvinconnections eliminate this drop with a pair ofleads added at the excitation terminals tomeasure and regulate the bridge voltage. For
example, when ie = 50 mA, RL = 5 Ω, and the
combined voltage drop in the two leads is
500 mV, no voltage drops in the sense wires.

Kelvin Strain-Gage Bridge Circuit
RL

i≅0

RL
R1

ie

R4
RL

R2

+
Vout Vexc
–
–
+

R3
RL
RL

i≅0
+
Vsense–ie

RL

Fig. 7.04. The Kelvin bridge circuit uses one pairof wires to provide the excitation voltage directly atthe bridge, and another to sense the excitation
voltage. A third pair of wires measures the bridge
output voltage. This arrangement removes thevoltage-drop error in the excitation wires from themeasured strain signal.

A commercial unit uses a Kelvin connection tomeasure and regulate the voltage at the bridge.
It supplies the voltage to the strain gage with
one pair of leads and measures it with anotherpair as shown in Figure 7.05. The six wires areused in pairs for Sense, Excite, and Measure.
The Sense lead is a feedback loop to ensurethat the Excite voltage is constantly held
within specifications.

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Data Acquisition Handbook

85

Commercial Strain-Gage Module+Sense

6

+Exc

5

–Exc
Optional
BridgeCompletion
Resistors

Excitation
Regulator–SenseInput Gain Adj.

4

+

3

–
R1

Exc. Adj.

R2

Shunt
Cal.

BypassBypassA

LPF

Scale Gain Adj.

To
A/D

A

OffsetAdj.

Note: Use R1 or R2, not both simultaneously.
2
1
Typical of 8 channels

Fig. 7.05. A commercial strain gage module provides adjustable excitation, gain, and offset for each channel.
This lets it make use of the instrument’s entire dynamic range.

Figure 7.05 (5.05)
Strain Gage Signal Conditioning

Most strain gage-based transducers and loadcells are assigned units of measure for weight,
force, tension, pressure, torque, and deflection
with a full-scale value measured in mV/V ofexcitation. For example, a load cell with a 10 V
excitation supply and a 2 mV/V gain factor
generates an output of 20 mV at full load,
whether the load cell was designed to handle 10,100, or 1,000 lbs. The difference is in the
resolution of the system. That is, the small 10 lb
load cell produces 0.5 lbs/mV, and the large
1,000 lb load cell produces 50 lbs/mV.

Measurement Computing(508) 946-5100

Conductors carrying such low level signals are
susceptible to noise interference and should beshielded. Low-pass filters, differential voltage
measurements, and signal averaging are also
effective techniques for suppressing noiseinterference. Furthermore, instrumentation
amplifiers usually condition the extremely low
strain gage signals before feeding them to
ADCs. For example, a 10 V full-scale input
provides 156 µV of resolution for a 16-bit
ADC. The amplifier gain should be adjusted toprovide the full-scale output of the strain gageor load cell over the entire range of the ADC.

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86

CHAPTER 7 Strain Measurements

Force and pressure transducers typicallygenerate an offset output signal when no
external force is applied. Instrumentation
amplifiers usually contain a control to adjust
this offset to zero and let the load cell cover
the full range of the ADC. Most instruments
also provide adjustable excitation and gain.

Wheatstone-Bridge Circuit
Strain gage
347 ΩCommon Mode Rejection Ratio

A high common mode rejection ratio (CMRR)
is essential for strain gage amplifiers. (See
Chapter 5.) CMRR is a measure of how well
the amplifier rejects common-mode voltages.
A strain-gage signal in a Wheatstone bridge issuperimposed on a common-mode voltage
equal to half the excitation voltage. For
example, consider a 10 V excitation supply(Vmax = 5 V) for a strain gage with 2 mV/V (Vs= 20 mV) at full scale and an amplifier with a
CMRR of 90 dB. (See Figure 7.06.) The amplifier can introduce 0.158 mV of error, corresponding to about 0.80% full scale, which
may not be acceptable:
Figure 7.06 (5.06)
EQN: 7.04. Common-Mode Rejection RatiodB = 20 log10(Vs/Ve)
Vmax/Ve = log10-1 (dB/20)= log10-1 (90/20)= 31,622
Ve = Vmax/log10-1 (dB/20)= 5.00/31,622= 0.158 mV% error = (Ve /Vs)100= (0.158/20)100= 0.79%Where:Ve = error voltage, 0.158 mV
Vs = signal voltage, 20 mV
Vmax = maximum voltage, 5V
CMRR = 90 dB
By comparison, a CMRR of 115 dB introduces
only 9 µV of error, which corresponds to only
0.04% of full scale.

Measurement Computing(508) 946-5100

+V

350 Ω

5.02 V
350 Ω

+10 V

20 mV

IA

Gain = 200
Vout = 200 (0.02)= 4.00 V

350 Ω 5.00 V –V

Fig. 7.06. Normally, the strain gage signal in a
Wheatstone bridge is superimposed on a commonmode voltage equal to half the excitation voltage.
Consequently, a high CMRR is necessary to reject
the common-mode voltage and amplify the strain
gage signal.

Strain gage signal-conditioning modules
usually provide a regulated excitation sourcewith optional Kelvin excitation. Onboard
bridge-completion resistors may be connected
for quarter and half-bridge strain gages.
Instrumentation amplifiers provide input andscaling gain adjustments, and an offset adjustment nulls large quiescent loads. This lets
input signals use the full range of the dataacquisition system and the measurements
cover the full resolution of the ADC.
Some strain gage signal conditioners provide
fixed gain, offset, and excitation settings, but
fixed settings do not take advantage of the
maximum dynamic range of the ADC. They
decreases the actual available resolution of themeasurement. For example, many genericstrain gage signal conditioner modules can be
set to a fixed 3 mV/V rating. At 10 V, the excitation, offset, and gain trimming are all fixedand no adjustments can be made.

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Data Acquisition Handbook

An excitation adjustment lets users set theexcitation voltage to the maximum allowed by
the manufacturer, which maximizes the bridge’s
output. Also, the offset adjustment lets users
zero the output offset produced by either a small
bridge imbalance or a quiescent deformation of
the mechanical member. And the gain
adjustment lets users set a gain that provides afull-scale output under maximum load, which
optimizes the dynamic range of the ADC.

CALIBRATION APPROACHESThe signal-conditioning module also typicallyprovides a shunt calibration feature. (See Figure
7.07.) It lets users switch their own shunt resistors
into either one of the two lower legs of the bridge
under software control. For example, a shunt
resistor can be calculated to simulate a full load.
Applying a shunt resistor is a convenient way
to simulate an imbalance without having toapply a physical load. For any balanced bridge,
a specific resistor can be connected in parallelwith one of the four bridge elements to obtaina predictable imbalance and output voltage.

87

For example, a 350 Ω, 2 mV/V strain gage delivers
full output when one leg drops by 0.8% to347.2 Ω. A 43.75 kΩ resistor shunted across one
or the other lower bridge elements swings theoutput to full positive or full negative.
An equation for calculating the shunt calibration resistor value is:
EQN. 7.05. Shunt Calibration Resistor for Transducers

Rs = Rba[Vex/4(Vo)]Rs = 350[10/4(0.020)] = 43.75 k
Where:
Rs = shunt resistor, ΩRba = bridge arm resistor, Ω
Rs >> Rba
Vex = excitation voltage, V
Vo = bridge output voltage, V
Many products include calibration softwarewith a Windows-based program that provides
several calibration methods, online instruction, and a diagnostic screen for testing thecalibrated system.

APPLICATIONS
Shunt Bridge Circuit
Ron
350 Ω

Transducers and Load Cells

+10 V

350 Ω
(+)

Rs =
43.75 kΩ

350 Ω

Close
to shunt350 Ω

Voltage ≅ 20 mV
when shunted

RTN
(–)

Fig. 7.07. A bridge may be calibrated with a shunt
resistor, which is switched with a software command
into either one of the two lower legs.

Measurement Computing(508) 946-5100

Strain gages are commercially available in prefabricated modules such as load cells that measureforce, tension, compression, and torque. Loadcells typically use a full-bridge configuration andcontain four leads for bridge excitation andmeasurement. The manufacturers provide calibration and accuracy information.

Strain Diaphragm Pressure Gages

A strained-diaphragm pressure gage consists of
two or four strain gages mounted on a thin
diaphragm. The gages are wired in a
Wheatstone bridge circuit, including bridge
completion resistors when needed, so the pressure gage is electrically equivalent to a loadcell. The output voltage is specified in mV/V ofexcitation for a full-scale pressure differential
across the diaphragm.

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CHAPTER 7 Strain Measurements

When one side of the diaphragm (called thereference pressure side) is open to the ambient
atmosphere, the gage compares the inlet pressure to the ambient pressure, which is about
14.7 psi at sea level. When the gage measuresambient pressure, the reference chamber must
be sealed with either a vacuum reference (nearzero psi) or the sea-level reference.
Temperature variations can affect the accuracyof these gages. A pressure gage with a sealed nonzero reference pressure exhibits temperature
variations consistent with the ideal gas law. For
example, a 5˚C change in ambient temperaturenear normal room temp (25˚C) produces anerror of 1.7% in the pressure ­measurement.
Temperature variations can also affect the performance of the strain gages themselves. Transducers

Measurement Computing(508) 946-5100

must contain ­temperature compensation circuits
to ­maintain accurate pressure measurements inenvironments with widely varying ­temperatures.
All strained-diaphragm pressure gages require aregulated excitation source. Some gages containinternal regulators, so users can connect anunregulated voltage from a power supply. Somestrained-diaphragm pressure gages also employ
internal signal conditioning, which amplifies
the mV signal output of the Wheatstone bridge
to a full-scale voltage from 5 to 10 V. Gages ofthis type have low-impedance outputs. In
contrast, other pressure gages have no internalsignal conditioning so their output impedance
equals the Wheatstone bridge resistance (severalkΩ for semiconductor types), and their full-scale
output is in mV.

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89

Chapter 8
VIBRATION AND SOUNDSENSORS FOR VIBRATION AND
SOUND MEASUREMENTS
Strain-Gage Accelerometers
Strain-gage accelerometers contain a small, calibrated mass attached to the free end of a cantilever beam. (Also see chapter 7.) One strain
gage sits on the top of the cantilever beam, andanother sits opposite it on the underside. Bothstrain gages measure the bend of the cantilever,
one in compression and the other in tension
when the device accelerates in a direction
perpendicular to the plane of the gages. First,
the product of the mass and acceleration
converts to force (F = ma) then force converts toan electrical signal. The gages are calibrated togenerate an output signal proportional to the

Accelerometer Frequency Response

amount of acceleration, typically measured ingravity units or gs. The accelerometers’ natural
frequency is usually in the order of several kHz,and in an undamped system, the cantilevermass combination forms a harmonic resonatorwith Q near 100,000.
A system with an extremely high Q excitednear its resonant frequency produces large
oscillations, which can mask the true acceleration signal, produce inaccurate measurements,
and damage the sensor. Consequently, oil or asimilar material is frequently added to control
the damping. Figures 8.01 and 8.02 show thefrequency response of harmonic resonatorswith Q = 1 and 100,000. For Q = 100,000, theaccelerometer is accurate within 10% to 1/3 ofits resonant frequency. In contrast, for Q = 1,
the accelerometer is accurate within 10% tohalf of its resonant frequency.

105

Low Frequency Response

104

Q=105

103

1.35

Amplitude

1.30
Amplitude

102
101
100
10-1
0.0

Q=1
0.5

1.0

ω/ω0

1.5

2.0

Fig. 8.01. An undamped accelerometer can have ahigh Q at its resonant frequency, which produces
large oscillations and completely overrides theacceleration signal. Also, if large enough, resonant
oscillations can destroy the accelerometer.

1.25

Q=105

1.20
1.15
1.10
1.05

1.00
0.95
0.0

Q=1
0.1

0.2

ω/ω0

0.3

0.4

0.5

Fig. 8.02. The low frequency portion of the signal
shown in Figure 8.01 is magnified to show its response.

.08)
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CHAPTER 8 Vibration and Sound

Most strain gages in accelerometers are wired
in Wheatstone bridge circuits. Signal conditioners for a common strain-gage bridge also
apply to this type of transducer since it
resembles a strain-gage circuit in a load cell.

time varying charge, Q(t). The charge is
proportional to the force on the crystal and isusually measured with either a voltage or
charge amplifier.

Piezoelectric Transducers

In Figure 8.03, the voltage amplifier’s gain is 1.
Adding feedback resistors or increasing thenumber of amplifier stages changes the gain.
The amplifier converts the high-impedance
voltage input to a low-impedance voltage
output. The voltage is Q/C, where Q is the
charge in coulombs, and the C is the capacitance in Farads, which includes both thesensor and the lead capacitance. When calibrating a charge amplifier, the cable capacitance is part of the charge equation, so
replacing the cable without recalibration canchange the instrument’s output accuracy.

Piezoelectric materials also are used in accelerometers and microphones. They generate electrical charges on opposite faces of the crystal
under dynamic mechanical forces including
compression, tension, and twisting. A varietyof transducers, such as microphones use piezoelectric elements to convert sound energy
picked up by a diaphragm into electricalsignals. Conversely, sonic transducers use
piezoelectric elements to convert electricalsignals into sound energy. Quartz is one of themost common materials applied in piezoelectric transducers and are readily available.
Another common material is a piezoceramicmaterial, composed of lead, zirconate, andtitinate (PZT).
From an electrical viewpoint, a piezoelectric
device resembles a capacitor containing a

Piezoelectric Voltage and Charge
Amplifier
Voltage amplifier
PZT

C

R

+
A
–

Voltage Amplifier

Charge Amplifier

The charge amplifier shown in Figure 8.03 is
extremely versatile because its output voltage
is V = Q/C where Q is the charge, and C is the
feedback capacitance. When the operational
amplifier has an extremely large open-loop
gain, the output voltage is independent of thecable capacitance. This lets the amplifier work
well even when it’s located several meters
away from the sensor. However, because
electrical noise increases in proportion to

Piezoelectric Sensor and Amplifier
BNC

AV
PZT

C

R

Charge amplifier–A+

Fig. 8.03. A high-gain op amp isolates the sensor’s
output voltage from the effect of signal cable capacitance and lets the amplifier reside several meters
away from the sensor.

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SensorFig. 8.04. Some piezoelectric accelerometers come withbuilt-in amplifiers. They have a low output impedance
and require an external power supply.

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Data Acquisition Handbook

cable length, noise susceptibility determines
the maximum permissible cable length morethan does device sensitivity. The charge
amplifier doubles as a high-pass filter withlower corner frequency f = 1/(2πRC), but thisrepresents a trade-off between sensitivity andfrequency response. Decreasing C increases
sensitivity, but it also increases the lower
corner frequency.
PZT-type sensors can’t make static measurements because of their leakage resistance andhigh output impedance, which ranges from 1010
to 1012 Ω. Such leakage resistance combinedwith a capacitance of several hundred pF, yields
a time constant of a few seconds.

Low-Impedance Transducers
Most modern piezoelectric transducers containintegrated signal conditioning amplifiers. (SeeFigure 8.04.) Also known as integrated-circuit
piezoelectric transducers, these units have a low
impedance output and require an external power
supply. The manufacturer usually specifies their
sensitivity and frequency range. Users need only

91

connect a supply to the power terminals and theoutput to a voltmeter circuit. The voltage is thenscaled to the measured engineering units.
The most common application for piezoelectric-type accelerometers is in measuring awide range of accelerations and mechanicalvibrations. They monitor automobile deceleration (and deploy the air bag at the correct
millisecond) in safety systems, lift-off acceleration and motion during space-shuttle
missions, and mechanical vibration innumerous machines. Piezoelectric sensors,
however, cannot measure constant acceleration as do strain-gage sensors.
Low-impedance piezoelectric transducers alsomeasure pressure or force. The accelerometercircuit requires only two wires to handle both
power and signals. Due to the sensor’s low
impedance, the system is not normally sensitive to externally introduced or triboelectriccable noise or cable length. Piezoelectric
sensors have resonant frequencies as high as
120 kHz giving them a usable frequency rangeof less than 1 Hz to more than 40 kHz.

Accelerometer and Input Signal Card
MOSFETS
G

BNC

AC coupled
CrystalR

FilterbypassC
PGA

D

Programmablefilter

SSH

BufferTo dataacquisition
systemProgrammable
clock

2 mA
Current
sourceTransducer

4 mA

Micro
controller

Controllines

Typical dynamic signal input card

Fig. 8.05. A piezoelectric accelerometer with a built-in MOSFET amplifier stage drives a programmable gain
amplifier on a dynamic signal input card. The 2 or 4 mA current source powers the MOSFET.

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CHAPTER 8 Vibration and SoundThe constant-current source provides asource-to-gate bias for the FET. As the gate
current responds to changes in applied pressure on the crystal, the drain-to-source voltage(Vds), and the voltage out of the preamplifierchange proportionally. An ac coupling circuit
or a high-pass filter is always necessary
because of the high dc offset at Vds thatdevelops from the bias current. The cutoff
frequency of the high-pass filter depends on
the application and the particular accelerometer. (See Figure 8.06.)

Figure 8.05 shows a simplified connection
scheme between an accelerometer and signalconditioning card. The voltage developed
across R is applied to the gate of the MOSFET,
which receives power from a constant-current
source of 2 to 4 mA. The MOSFET circuits bias
off at approximately 12 V in the quiescentstate. When the system is excited, voltage
develops across the crystal, which is applied
to the gate of the MOSFET. The voltage
produces linear variations in the MOSFET’s
impedance, which, in turn, produces a
proportional change in the bias voltage. This
voltage change couples to the input amplifier
through capacitor C. The value of R and theinternal capacitance of the piezoelectric
crystal control the low-frequency corner.
Units weighing only a few hundred grams
generate high-level outputs to 10 mV/g withresponse to frequencies from 0.3 Hz to 2 kHz.
Smaller units with less sensitivity respond tofrequencies from 1 Hz to 35 kHz.

Constant-Current Source

iL

R

R1

Load

–
+

To eliminate the need for an outboard preamplifier, some accelerometers contain the current
source and the ac coupling circuitry. Thisfeature also lets accelerometers connect to thedata acquisition system through simple BNC
connectors. Most accelerometers, however,require an amplifier and filter on the output
before the analog to digital conversion stage.
Also, the programmable-gain amplifier lets theoperator adjust the gain for optimum response.

+

–

VREF

+

–

A

R4

A

R3

R2

VREF

VREF

iL =

–

Floating load

iL

R

Load

+
If:

R3

R2

=

R4
R1

Then: iL = –Grounded load

VREF
R2

Fig. 8.06. A constant current source provides a source-gate bias for the field-effect transistor in the accelerometer. Depending on the specific FET circuit, a constant-current source may be configured for either a floatingload or a grounded load.

Figure 8.06 (5.14)
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Data Acquisition Handbook

Along with programmable amplifiers, programmable low-pass filters reject unwanted highfrequency signals. The signals typically comefrom noise or high-frequency vibrations that
do not relate to the application. When developing the front-end circuitry for this type of measurement, noise rejection and bandwidth are

Measurement Computing(508) 946-5100

93

primary concerns. As the bandwidth increases,
the noise can increase as well. Low-pass filters
also should be used in most accelerometer conditioning circuits to reduce noise and aliasing
effects. The cutoff frequency of the low-passfilter should be close to the system’s maximum
useful operating frequency.

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CHAPTER 8 Vibration and Sound

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95

Chapter 9
DISPLACEMENT AND POSITION SENSINGLINEAR DISPLACEMENT

Motion control systems typically require asensor in a feedback loop to ensure that themoving member or component reaches its
commanded position. Position sensors are
typically used on machine-tool controls, elevators, liquid-level assemblies, forklift trucks,
automobile throttle controls, and numerous
other applications. Position sensors measure
either absolute or incremental displacement,
and are made with a variety of different materials. Also, when the sensor’s power fails, someretain the measured position data, and othertypes lose the information.
One common device is a resistive position
sensor or potentiometer. It may be a singleturn or a multi-turn rotary sensor with anelement made of carbon or any one of anumber of more durable, high-temperature,
conductive-plastic
Some
types
Figure 9.01film
A materials.
Figure9.01
Bof potentiometers connect to the moving
member through a rigid or flexible coupling
while others are actuated with a string
wrapped around its output shaft. Yet otherposition sensors include LVDT’s (linear
variable differential transformers), sonicallyoperated devices, and digital encoders.

String-Potentiometer Transducers

String-type potentiometers are basically rotational devices that typically measure linear
motion as shown in Figure 9.01. Because a
Figure 9.01 C
cable connects between the moving member
and the potentiometer, the cable must be
preloaded with a spring, have a low coefficient

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String-Potentiometer Position Sensor
Wire wound

Film

Direction of
linear motion
Direction of rotation
String or cable

Coil spring
Potentiometer
Threaded
drum

Fig. 9.01. String potentiometers occupy relatively
little overall space in a machine installation or
device under test. In some applications where asensor cannot be mounted on the device being
measured, the potentiometer may be conveniently
located some distance from the test bed without
sacrificing accuracy or repeatability.

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CHAPTER 9 Displacement and Position Sensingof expansion under wide ranges of temperature (-65 to +125˚C), and have little tendency
to stretch. Because of these constraints, its
accuracy is somewhat less than that of a morerigid, shaft-coupled sensor.
Most modern potentiometers are extremely
linear and are made with special alloys or
precious metal wipers and conductive plastic
resistive elements to withstand as many as
100 million revolutions. They are usually part ofa voltage divider and provide an output voltage
proportional to displacement, but they may
also drive analog-to-digital converters to feed
microprocessor-based instruments directly.
However, when digital outputs are needed, amore common practice uses digital encoders
that inherently supply higher accuracy.

LVDT: Linear Variable Differential
Transformers

LVDTs are electronic devices comprised of a
special transformer with a movable metal actuator in its hollow-core. The transformer contains
two differentially wound secondary windings oneither side of a primary winding. As the actuatorin the core moves in and out, it couples a signal
from primary to secondary in proportion to thedisplacement. Being differentially wound, LVDTs
also can provide directional signals with theappropriate signal conditioning circuits. With
the actuator at the center position, the output
signal is null or zero. As the actuator moves out
in one direction, the signal is positive, and when
it moves in the opposite direction from null, thesignal is negative. (See Figure 9.02.)

LVDT Position Sensor
Physical Construction
S1

Excitation and Output Signals
Primary
excitation
Sec. 1
output

LVDT

P
S2Sec. 2
outputCore motion

Zero output at null

Winding Positions
Secondary 1

Primary coil

Secondary 2

Sec. 1 +Sec. 2

Schematic
Primary coilCore motion

Core motion

Figure 9.02 C
Secondary 1

Secondary 2

Fig. 9.02. LVDT sensors provide distance and direction information with extremely high accuracy. But a drawback for some installations is their relatively long length. The minimum length of the sensor is its body, andthe core can extend only as far as the length of the body.

Figure9.02 B Computing
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igure 9.03

Data Acquisition Handbook

LVDTs require electronic circuits to provide anac signal to excite the primary and a demodulator to convert the output to a dc signal withdirectional polarity. Some LVDTs contain theelectronics onboard and only require power toobtain an output, while others require separate signal conditioners or modules. The LVDTs
themselves are rugged devices and have a
MTBF of as much as 3 million hours. The main
limitation is the displacement range of thecore to ensure high linearity.

Magnetostrictive Position Sensor
Another relatively new type of position sensor
is based on the theory of operation of theinteraction of two magnetic fields on magnetostrictive materials. The major componentsof the sensor include an external permanent

97

magnet, integral sensing and output signal
conditioning circuits, a sensor tube, and awave-guide element. The permanent magnet
fastens to the movable object, and the housing
remains attached to a stationary reference
point. (See Figure 9.03.)
The output signal conditioning circuit generates a current pulse and magnetic field which
travel down the waveguide. When the transmitted field reaches the permanent magnet
field, the combination induces a strain pulsein the magnetostrictive tube. The strain pulse,
in turn, travels down the waveguide to thehead of the sensor where it is detected. The
time measured between the application of theinterrogation pulse and the arrival of thestrain pulse determines the precise, absolute
distance between the movable magnet and theMagnetostrictive Position Sensor

Interrogation:• Return wire
• Wave guide

The magnetic field generated
by the interrogation pulse
encompasses the entire
waveguideInteraction of magnetic fields
causes wave guide to generate
a strain pulse

Protective sheath

Pulse
outputEnd cap
Magnetic field
generated byposition magnet

Strain pulse
detector

Wave guide
Movable position magnet

Distance

Fig. 9.03. Magnetostrictive position sensors have a high signal-to-noise ratio, shock and vibration resistance,
and accuracy. They are rugged devices that deliver from 30 to 400 mV signals at their output terminals andrequire little or no additional signal conditioning. A possible disadvantage for some applications, however, is
its wave-guide, which must be at least as long as the measured stroke.

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CHAPTER 9 Displacement and Position Sensing

pulse generator. Resolution is about 0.002
mm, durability is high, and the stroke ranges
from 5 to 20,000 mm. Higher resolutions
require additional interpolating circuits. The
output signal, which is proportional todisplacement, can be either analog or digital,
and it can represent both position and velocity.

Linear Encoders

Linear encoders have evolved over the years
to provide extremely accurate position information. Their principle of operation depends
upon sensing position from precision graduations (position codes) etched on a linear
scale. They come in both contacting andnon-contacting types, and their length isprimarily limited to their temperature-sensitive coefficient of expansion to preserve accuracy. Most modern scales are glass, and opticalsensors detect the graduations. (See Figure
9.04.) The benefit of a non-contacting type istypically longer life because it uses optical

Two-Channel Linear Position Encoder
Stationary plate
number 1
Movingplate
Directionof motion

ROTATIONAL POSITION

Rotational position may be measured withpotentiometers, coupled to a rack and pinion
assembly or gear trains to obtain a suitable ratioof potentiometer revolutions to measured
component rotation or movement (see figure9.05). Potentiometers commonly come in single
turn to as many as 25 turns and fit numerous
applications. But, most rotational measurements, whether they are velocity, rpm, or position, are best handled with digital encoders.
Like linear encoders, several types have beenused over the years including both contacting(see Figure 9.06) and non-contacting. They
contain a disc (code wheel) or a plate made withalternating opaque and transparent segmentsthat move between a light source such as anLED, and a photo-detector. (See Figure 9.07.)

Rack and Pinion Encoder AssemblyLight sensor number 1
Light sensor number 2

Stationary plate
number 2 (180˚ out ofphase with number 1)

Lens
Light
sourcecomponents instead of brushes or a contacting
wiper. All optical encoders contain a lightsource, light detector, code wheel or scale,and signal processor.

Encoder
Adjustable springloaded mountto accommodate runout

Light
rays

Alternating opaque and transparent
lines interrupt light from source to sensors

Fig. 9.04. Two-channel encoders provide step anddirection vectors with an output signal that can be asinewave, squarewave, or a series of equally spaced
pulses generated at regular intervals on the waveform.

Figure 9.05

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Fig. 9.05. A rack and pinion sensor assembly shouldhave a flexible mount to compensate the combined
runout error. In a machine application, the sensor
should be protected from metal chips and other foreign
objects to ensure measurement accuracy.

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Data Acquisition Handbook

Absolute Contact Encoder Disc
20
21
22
23

4-bit binary
word

B0

Shaft

B1
B2
B3

14
13
12
11
10

9

15 0 1

2

.

8

7

6

5

3
4

99

Absolute Encoders

Absolute encoders typically contain several
detectors and slots on a revolving wheel toprovide a unique output binary code for each
shaft position within its range of resolution.
Therefore, the shaft position can be determined absolutely. Moreover, the correct position is always retained before and after a
power failure. The tracks on the encoder
wheel may be arranged to provide a 0.5 bit
maximum error. Absolute encoders come insingle and multi-turn versions.

Incremental Encoders

Contact
tracksFig. 9.06. The disc pattern on the contacting encoder
is a digital word, which represents the absolute position of the encoder shaft.

Incremental encoders usually provide twochannels of square wave output signals. Eachpulse represents an increment of rotation ofrelative position. These encoders often containa third slot called the reference slot used tocalibrate the encoder. Dual channel encoders
are more accurate than single channel types.
They register direction, and when a code
wheel stops on the edge of a slot, incident
vibrations cannot generate an error count, as
it does for single-channel sensors.

Rotary Position Encoder
L.E.D. light source

Optical disk

PC board

Detector

ReticleCollimated light
Signal andpower

Fig. 9.07. The shutter is one of the most critical components of the encoder assembly. The sensor’s resolution
and accuracy hinge on the quality of the vendor’s manufacturing and assembly processes.

Figure 9.07
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CHAPTER 9 Displacement and Position Sensing

Quadrature Encoders

The two channels, A and B, of a dual output
incremental encoder are usually displaced by
90 degrees from each other. Therefore, thespacing between pulses is one-half the line
width. When the disc contains 1024 lines and1024 spaces, the sensor generates 4 x 1024 =
4096 pulses per revolution; hence the output
signals are delivered in quadrature. (See Figure
9.08.) The angular movement in this arrangement resolves to 5.27 minutes of arc.

Two-Channel Quadrature Output
Channel A
Channel B
eMAX
p

p

p

p

p

p

eMIN

p

p

p

Plate
movement

One line width displacement

Fig. 9.08. The two light sensors in a dual channel
linear position encoder are connected in a push-pull
configuration and track one another, so equal changes
in the input produce equal changes in the output.

Signal conditioning circuits generate theoutput signal waveform and complementarysignals, sense their direction, and filter the
output. The amplitude of the input voltage
controls two stable output states of a shaper
circuit. The shaper output feeds several gates,
which in turn generate the pulses that markthe encoder’s zero-signal crossings. The circuits
also sense whether channel A leads or lags
channel B to provide direction information.

of instrument and quantization errors, as wellas manufacturing and assembly errors. For
example, the light sources must be closely
matched and the code wheel must have low
disc eccentricity.
Encoder error is relatively easy to quantify,
whereas total error depends upon each application and is a little more difficult to pin down.
Several sources contribute to total error, including manufacturing and assembly tolerances,
LED characteristics, scale and disc alignment,
and power supply stability. Others include electrical interference, temperature variations,mechanical coupling, and mechanical vibration.
In some cases, encoder misalignment errors
can be calculated. For example, a linear sensor
often incorporates a glass scale to sense linear
displacement. When the scale is bent orbowed, the curvature introduces an error thatis proportional to the displacement d, as
shown in Figure 9.09. When d is small relative
to S, the concave surface of the scale is shortened by ∆, and the convex surface is elongatedby ∆, where ∆ = 4td/S.
For example, when:S = 8 in.
t = 0.20 in.
d = 0.0010 in,then:∆ = (4 x 0.20 x 0.0010)/8 = 0.0001 in.
Lines etched on the convex side produce anerror of +0.0001 in., and lines on the opposite
side produce an error of -0.0001 in./foot.

ErrorsPower supply stability and freedom from electrical interference are critical requirements forencoders. Both anomalies can produce intermittent and random operation, which arefrequently difficult to diagnose.

Encoder manufacturers endeavor to minimize
device errors, but users must be concernedwith total system errors. Encoder errors consist

Temperature has minimal influence on today’s
encoder electronics, although the maximum
operating temperature at rated power for LEDs

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Data Acquisition Handbook

Misalignment Error in Glass Scales
Top view ofglass scale

S

d

t

Fig. 9.09. The top view of an exaggerated bow inlinear scale illustrates the errors that could undermine measurement accuracy. The error in this caseis proportional to the displacement “d.”Measurement Computing(508) 946-5100

101

is 150˚F compared to 200˚F for incandescentlamps. But LEDs have more than double theexpected life of lamps, and they work withsuperior signal conditioning circuits, so they
are used almost exclusively. Temperature fluctuations affect linear glass scales more significantly. For example, a ±150˚F temperaturechange can produce an error of ±0.0008 in./ft.
The mechanical coupling to the moving
member should suit the application, considering axial and radial loads, acceleration,
velocity, vibration, accuracy, and resolution.
The proper coupling reduces errors andincreases the encoder’s life.

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CHAPTER 9 Displacement and Position Sensing

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103

Chapter 10
NOISE REDUCTION AND ISOLATIONCONTROLLING NOISE

Controlling noise in measurement systems isvital because it can become a serious problem
even in the best instruments and dataacquisition hardware. Most laboratories andindustrial environments contain abundant
electrical-noise sources, including ac powerlines, heavy machinery, radio and TV stations,and a variety of electronic equipment. Radiostations generate high-frequency noise, while
computers and other electronic equipment
generate noise in all frequency ranges.
Building a completely noise-free environment
just for running tests and measurements isseldom a practical solution. Fortunately,
simple devices and techniques such as using
proper grounding methods, shielded andtwisted wires, signal averaging methods,
filters, and differential input voltage amplifiers
can control the noise in most measurements.
Some techniques prevent noise from entering
the system, while others remove extraneous
noise from the signal.

behaves in various types of circuits. For example,
high redundancy is one key feature that makes
most of the electrical distribution systems
around the world safe and operate properly.

Grounding for Safety

Isolated secondaries of step-down power distribution transformers are generally groundednear the transformer and within the firstswitching panel in the wired path to the eventual load. (See Figure 10.01.) The ground is apoint within the panel connected to a nearby
earth ground rod. Typically, a large or significant structure (building frame) or metallic
system (plumbing) is also connected to theControl Panel Grounding
Utility
distribution
transformer

Circuit breakerbox

Controlpanel
Hot phase A
Hot phase B

THE GROUNDING CONFLICTA non-technical dictionary defines the term
ground as a place in contact with the earth, acommon return in an electrical circuit, and an
arbitrary point of zero voltage potential.
Grounding, or connecting some part of an electrical circuit to ground ensures safety for
personnel and it usually improves circuit operation. Unfortunately, a safe environment and arobust ground system often do not happen
simultaneously. It takesplanning
Figure10.01 based on
systematically understanding how electricity

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NeutralGroundEarthgroundEarthgroundEnclosuregroundFig. 10.01. Utility power transformers are typicallygrounded to earth ground near the transformer, andagain at the input to the electrical junction box or
first switching panel. The switching panel, in turn,
may be connected to a rod driven into the earth toensure that it too is at true ground potential.

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CHAPTER 10 Noise Reduction and Isolation

same point. This minimizes voltage differencesthat may develop between a water pipe and an
appliance with a three-wire grounded cord, forexample. An electrical fault such as a nongrounded conductor contacting a metal object
is designed to open a fuse or trip a breaker rather
than leave an electrically energized appliance ata higher potential than a nearby water pipe or asink faucet. If the ground connection in thepanel disconnects for any reason, the redundant ground near the transformer will provide
the path for fault currents to open fuses or trip
breakers. Preventing electrical shocks and electrical fires is the highest priority for groundcircuits, but the redundancies built into manyelectrical grounding systems occasionally limit
certain kinds of connections for input to dataacquisition systems.

Grounding for Robust Instrumentation

Several internal, common busses in a data
control instrument are arranged to regulate
current flows and terminate all paths at one
common point. This approachensures that
Figure 10.02
the current flowing in any path will not force
a voltage drop in a return path for anothercircuit and appear as an (erroneous) input
signal. (See Figure 10.02.) Usually, this one
common point connects through low impedances to the safety ground connection on the
instrument’s ac power cord. This connection
prevents the internal system from floating at
an ac potential between earth ground and the
input ac supply potential.

GROUND LOOPS

Measuring instruments that contain an earth
ground as described above usually generate aground loop. A ground loop can become aserious problem even when the ground voltage
on the measured point equals the groundvoltage entering the instrument through theline cord. A voltage that develops between thetwo grounds can be either an ac or a dc voltage
of any value and frequency, and as the voltage

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Common Grounds
VCC

VS

DigitalcircuitsPower
supplyADC

VSS VDD
Analogcircuits

DigitalgroundAnaloggroundPreferredcommon returnDefault groundEarthgroundEnclosuregroundFig. 10.02. Significant current fed to multiple circuits should have individual return paths to a common ground or negative terminal. This reduces therisk of voltage drops developing over long groundruns or lead wires that could become input (error)
signals to other circuits.

and frequency increase, the effects of theground loop become more troublesome.

Dangerous and Destructive Ground Loops
A transient current can generate a substantial
voltage on grounded conductors. During anelectrical fault when an energized conductor
contacts a safety ground, for example, a fraction of the supply voltage can end up on thesafety ground before the fuse or circuit breakersupplying the fault opens and removes thevoltage. This happens in a few milliseconds
and is usually not a safety hazard. But the
problem can be much more serious if lightning strikes a safety ground structure andthousands of amperes flow through the groundsystem. Potential differences across even afraction of an ohm can easily exceed 1,000 Vac
and damage equipment and endanger lives.

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Data Acquisition Handbook

Symptoms of Ground Loops

Sometimes, a measurement error is mistakenly
attributed to a ground loop problem, especially where a ground is not strictly involved.
The phenomenon relates to two types of situations; shared current flow in a circuit path,
which produces unintended voltages, andinadvertent circuits that interfere with theproper operation of intended circuits.

How Ground Loops are Created

A ground loop problem can be illustrated by
the following example. An integrated sensor
with internal signal conditioning contains
three wires; a positive power supply lead, asignal output lead, and a negative lead thatserves as both the power return and signal
common. (See Figure 10.03.) The sensor’s
internal circuitry draws about 30 mA and theoutput signal ranges from 0 to 5 Vdc.
The sensor is stimulated and a digital voltmeter reads the correct output of 2.50 Vdc on
the test bench. But when the three leads areextended by 500 feet with 20 AWG wire (10.4
Ω/1,000 ft at 20˚C), the common lead wire
carrying the 30 mA of power supply current
drops about 150 mV. This lead resistancevoltage drop adds to the sensors output voltage
and delivers 2.65 Vdc to the digital voltmeter.
The error amounts to about 6.6% and what’s
Figure 10.03
worse, it varies widely with the temperature of
the wire. The specific application determines
whether the error can be tolerated or not.
When the current drawn by the sensor circuit
is not a steady state value, that is, it consists ofan ambient level combined with a dynamic
component, the error introduced will be timevarying. It might be a relatively high frequency,
which acts as noise in the measured output, or
it may be in perfect synch with the physicalphenomenon being sensed. It then affects themagnitude of the time-varying output signal.
Both types of errors often appear in dataacquisition systems.

Measurement Computing(508) 946-5100

How to Eliminate Ground Loops

A reliable trouble-shooting method analyzes
current flow and predicts its results. The wires
from the intended point of measurement mustcarry only current associated with the bias
requirements of the analog input channel. (SeeFigure 10.04.) These currents are typicallymeasured in microamperes. At lower voltage

Interfering Ground loop
Sensor
+
+
Vout

30 mA

RL
–
+
150 mVVin = 2.65 V

Vout = 2.50 V
–

+

A

–

ES

150 mV+–

30 mA RL
500 ftFig. 10.03. The 150 mV dropped across RL in thebottom ground return line arises from the 30 mA ofcurrent flowing in 5.2 Ω of lead wire resistance. The
voltage adds to the sensor’s 2.50-V signal to yield
2.65 V at the input to the signal-conditioning amplifier and produces a 6.6% error.

levels, they can be altered substantially if forced
to share extremely long wires carrying merely
mA. A detailed wiring diagram and a circuit
schematic can provide insight and understanding to help prevent this type of problem
before hundreds of feet of wire are installed.
Frequently, multiple wires running between two
locations cannot be shared. When a common
wire is shared, the current in one channel affects
the voltage reading in another channel. In theprevious numerical example, a fourth wire
connected to the lower end of a differential
measurement channel provides an output

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CHAPTER 10 Noise Reduction and Isolation

Static Crosstalk

Bypassed Ground Loop
Sensor

150 mV
–
+

30 mA
+

Vout =
2.50 VVin =2.50 V+RL

150 mV

–

A

–

ES

30 mA

500 ftFig. 10.04. A separate wire run from the sensor
ground (or common terminal) bypasses the power
supply ground wire voltage drop. The true output
signal of the sensor (Vout) reaches the amplifier input
terminals because the input draws negligible current.

voltage that can be measured accurately with ahigh degree of confidence. This approach ismost effective when the system supports threewires, and they share a common power supply.
Differential input connections used with theanalog common, which is referenced to thepower supply return terminal, eliminates theeffect of the ground loops inherent in thismultiple-sensor arrangement.

CROSSTALK IN DATA ACQUISITIONSYSTEMS

Another type of ground-loop error is crosstalk
between channels. This may be defined as aninteraction between readings on two or morechannels, which may be static or dynamic.
When multiple channels are used and groundloops exist, the simplified errors described
previously most likely will be compounded bycontributions from other channels. The crosstalk
may or may not be obvious.

Measurement Computing(508) 946-5100

Consider a group of static channels with steady
voltages, that when measured individually,yield accurate readings. However, when eachchannel is connected to an input of the dataacquisition system and the readings change,
the change indicates that crosstalk is generated
by a steady-state ground loop. Likewise, when
the reading of a channel changes by connecting
another channel, crosstalk exists and the
problem is a ground loop.

Dynamic Crosstalk

Dynamic crosstalk is the name given to thesituation where a known dynamic signal on aparticular channel appears in a physically unrelated channel. The steady-state currents drawn
by the transducers discussed in the previous
example are idealized for simplicity. These
currents commonly vary with the measured
physical variables along with the errors.
Sequentially reading signals of widely varying
magnitudes produce sequential crosstalk in
multiplexed data acquisition systems.
Capacitive or inductive coupling between
channels generates crosstalk in systems withimproperly or carelessly dressed wires.
Generally, however, these are not attributed to
ground loops and are less common.

SHIELDED WIRING
Benefits

Metallic shields placed around equipment andtest leads effectively prevent noise from either
entering or leaving the system. For example,
loose or exposed wires become antennas forradio frequency signal pickup, and can form
loops that radiate noise.
To emphasize the need for controlling noise,
Figure 10.05 shows a single-ended voltage measurement on a shorted channel. Approximately
6 feet of wire, not twisted or shielded, wasattached to the data acquisition system. Figure

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107

Data Acquisition Handbook

10.06 shows the noise in a single-ended, shorted
channel using shielded cable with obvious
improvement.

Proximity to other wires, especially powerwires carrying high voltages and high currents
can couple noise into low-level signal conductors. Capacitive coupling can exist between
any two pieces of metal in close proximity,
including two conductors in totally separate
circuits. Likewise, air-core transformer coupling
can crop up between two closed wiring loops
Figure 10.06 (7.02)
in totally separate circuits.

Single-ended Measurement:
Unprotected Wires
5.00E-03
4.00E-03

Volts

3.00E-03
2.00E-03

1.00E-03
0.00E+00-1.00E-03-2.00E-03-3.00E-03-4.00E-03-5.00E-03

0

0.05

0.1
0.15
Time [seconds]

0.2

0.25

Fig. 10.05. Radiated noise spikes are easily picked up
on non-twisted, non-parallel, unshielded lead wires
connected to the single-ended amplifier input terminals, even when the inputs are shorted together.

re 10.05
Measurement Computing(508) 946-5100

5.00E-03
4.00E-03
3.00E-03
2.00E-03

Volts

The best instrumentation wiring schemes
consist of carefully grouped lines, twisted inpairs, occasionally covered with a second
shield, and routed through a dedicated conduit
or raceway. A shielded, twisted pair is quite
commonly used in a channel to connect asignal from a source to an input terminal.
Shields minimize capacitive coupling andtwisted wires minimize inductive coupling.

Single-ended Measurement:
Shielded Wires

1.00E-03
0.00E+00-1.00E-03-2.00E-03-3.00E-03-4.00E-03-5.00E-03

0

0.05

0.1
0.15
Time [seconds]

0.2

0.25

Fig. 10.06. Shielded wires reduce electrical noisepick up at the input of single-ended amplifier
input terminals, but other techniques often are
more effective.

Proper Installation and Use of Shields

Typically, a shield terminates at one end only,
unless it extends to the shield in another spanof the same channel wiring. The shield canterminate at either the transducer end or the
input channel end, but not both. When thesensor or transducer is in a shielded metallic
enclosure, which is also connected to earth
ground, the shield may be connected at thesensor end and remain open at the input
channel terminals. When the sensor is wellinsulated, the shield may float and connect tothe analog common of the data acquisition
system input terminals. Occasionally,
multiple-conductor cables composed of abundle of wires and an overall shield are
acceptable for a group of high-level, dc or lowfrequency signals, but would not be recommended for the general data acquisition case.
Compromising a well planned wiring systemwith low quality wire, shared conductors or
shields, and parallel, untwisted wires willproduce less than optimum results.

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CHAPTER 10 Noise Reduction and Isolation

ISOLATION AND FLOATING DATAACQUISITION SYSTEMS
Isolation

Isolation is defined as the separation of one
signal from another to prevent unintentional
interaction between them. All multiplexed
data acquisition systems contain a certain
degree of channel-to-channel isolation; relaybased systems have galvanic isolation whilesolid-state systems do not. Galvanic isolationis the absence of any dc path. Most isolation
methods eliminate all dc paths below 100 MΩ.
Three major benefits of galvanic isolation are
circuit protection, noise reduction, and highcommon-mode voltage rejection, especiallythose developed by ground loops.
Computer-based data acquisition equipment
makes possible an array of multiple channel
measurements previously beyond theeconomic reach of many applications. Thishas been accomplished by user acceptance of
two major compromises, multiplexing andnon-isolated inputs. Multiplexing is successfulwhen the sampling rate is adequately highand the source impedances are sufficiently
low. Lack of isolation places an entirely
different kind of limitation on the type of
input signals that can be connected.

Circuit Protection

Isolation separates the signal source from themeasurement circuitry that could be damagedby the signal. Voltages higher than about
10 V can distort data or damage components
used in the system. High-voltage input signals
or signals containing high-voltage spikes
should therefore be isolated. The protection
also works in the opposite direction to safeguard a sensitive signal conditioner from adevice failing elsewhere in the system.
Computer-based data acquisition equipment
is most often connected to a host computer,which is connected to earth ground. The

Measurement Computing(508) 946-5100

analog inputs of plug-in cards and mosteconomical external systems are not electrically isolated from earth ground or eachother. Many applications are compatible withthis situation, but some applications face a
problem with high common-mode voltage.

Rejection of High Common-Mode Voltage

Common-mode input voltage is defined as
the voltage applied between the common
terminal and the two input terminals withthe condition that the two input voltages be
identical. In other words, the two input
terminals may be connected together and thecommon-mode voltage applied between the
shorted inputs and the common terminal as
shown in Figure 10.07. In a practical test andmeasurement situation, the common-mode
voltage may exceed the instrument amplifier’s input rating, which is typically less than
10 V. For safe and accurate measurements,
common-mode voltages higher than 10 V
must be isolated from the instrumentation
amplifier while allowing the measured signal
to pass. Common types of isolation amplifiers use magnetic, optical, or capacitivemeans to couple the signal.

Magnetic Isolation

Special instrument amplifiers use transformers that magnetically couple analog-type
ac signals from the input section to the
output section while effectively sustaining
high common-mode voltages. Transformer
coupling also lets them provide isolated
power to the input stage without using aseparate dc/dc converter. A particular instrument amplifier contains an input op amp
with a CMRR of about 130 dB at a gain of100, and 2000 V peak common mode voltage
isolation. Similar instrumentation amplifiers
are available for powering isolated bridges,
cold junction compensation, linearization,and other special signal-conditioning requirements. (See Figure 10.08.)

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Data Acquisition Handbook

Optical Isolation

Common-Mode Input Voltage
Rf
100 kΩ
100 kΩ–

Ri
100 kΩ

+
VCM
–

+

Ri

A
VoCM =ACM VCM

95.3 kΩRf0.1% resistors

Common-mode
adjustment

109

Fig. 10.07. Common-mode voltage is measured
between the two input terminals and the common
terminal. Because the two inputs must have identicalvoltages, they may be tied together and connected to
one voltage source.

Optical isolation is now the most commonlyused method to couple digital signals. The
measured input voltage signal is converted to a
current, which activates a light-emitting diodewithin an optical coupler. A light-sensitive transistor located adjacent to the diode, but on the
opposite side of a voltage barrier, converts thelight signal back to a current that the instrumentation amplifier can handle. The voltage barrier
typically provides as much as several thousand
volts of isolation between input and output.
Optical devices also are commonly used toisolate the output of an ADC, which is usuallya serial string of data pulses passing through asingle optical coupler. (See Figure 10.09.) The
serial string is often converted from several
parallel signals (ranging from 8 to 24 output
ports, for example) to minimize the number of
optical devices required in a system. Parallel toserial conversion circuits are less expensive
than the 8 to 24 optical devices (one for each
bit of output from the parallel ADC). In theseTransformer Isolation Technique
Input

IN–

IN+
Shunt

VRS= 5 V

IN COM+Viso Out
–Viso Out–+

OutputA

Modulator+

SignalDemodulator

HI
VOUT = 5 V
LO

–

Power
Rectifier& filter

Oscillator+15 V DCPower returnFig. 10.08. This instrumentation amplifier with transformer coupling is typically used in an industrial process control
loop and provides galvanic isolation in both input and output circuits. It can measure a ±5 Vdc signal riding on acommon-mode voltage as high as 2000 V.

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110

CHAPTER 10 Noise Reduction and Isolation

Optical Isolation Technique
Analog section
A/D

AC
line

Isolated
power

A

Opto

A

T/Cs

Opto's Control
signals

Digital section

0.1 µF

Figure 10.11 illustrates a typical multiplechannel, programmable isolation amplifier ina data acquisition system that uses all three
types of isolation: transformers, opticaldevices, and capacitors. A transformer-based
dc/dc converter supplies power to the isolatedside. A capacitively coupled device isolates theanalog signal while two optical couplerstransmit the digital control signals to thefloating circuitry.

Fundamental Application Mistakes
10 Ω

Fig. 10.09. Optical devices are the most widely usedcomponent for coupling signals across high-potential
barriers and for low-level signals that are prone tothe electrical noise typically coupled through different
ground potentials.

instances, the power supply for the ADC andassociated input circuitry are also isolated,usually with a transformer.

Dangers in measurement can easily surface
when a non-isolated analog input to a dataacquisition system is erroneously connected to a
device operating at high common-mode voltage
with respect to earth ground. The mistake,which often precedes this problem, is determining the voltage range of the desired signal
with a handheld digital or analog multimeter,
and neglecting the relationship of the signal

Modulator/Demodulator Coupling
Input

Capacitive Isolation

A capacitor is a passive device that couples ac
voltage from one stage to another while
blocking the dc component. By this definition,it’s a simple but inexpensive isolator. The
measured signal to be isolated is modulated
and coupled through the capacitor to thereceiving side. On the receiving side, the ac
signal is demodulated to restore the original
signal. This technique is often applied to
low-cost isolation amplifiers where thecoupling capacitor is composed of a common
layer between two isolated IC substratesections. Signal isolation using these
specialized ICs is rated as high as 1,500 V. The
main benefits of this approach are simplicity,
low cost, and bandwidths as high as 50 kHz.
Figure 10.10 illustrates a dc/dc converter
often used as the modulator/demodulator in
an isolation amplifier.

+VC
Vin
-VC
Com 1
Gnd 1
+VCC1
Ps Gnd-VCC1

OutputSense

C1
Duty cycle
modulator
Sync

Duty cycle
demodulator
C2
T1

Rectifiers
Filters

Oscillator
driver

Vout
Com 2
-VCC2
Sync
+VCC2
EnableGnd 2Isolation

Fig. 10.10. Low-cost isolation amplifiers couple dc
signals by modulating the signal, coupling its ac
equivalent through a substrate capacitor connecting
the stages, and then demodulating the signal torestore it to its original dc value.

Figure 10.10 (5.22 new book)Measurement Computing(508) 946-5100

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Data Acquisition Handbook

Isolation in Motor Drives

Programmable Amplifier
Barrier
Opto couplers

Isolation transformer

Configuration
µP
Hi
Lo

Isolation
amplifier
Programmableattenuator
amplifier

Motor
drive

Speed adjustment potentiometer

LPF

M
U
X

To A/D

Typical of each channel

Fig. 10.11 All three techniques of isolation are used
in a multi-channel, programmable amplifier. Opticaldevices couple digital signals, capacitors couple analog signals, and the dc/dc converter supplies power tothe isolated side.

Figure 10.12

with respect to earth ground. If connected withthe dotted lines, as shown in Figure 10.12, themotor drive can be damaged as soon as power isapplied. This is because the control circuitry inmany ac and dc motor drives is necessarily referenced to high voltage with respect to earth
ground. When motor current is sensed with ashunt or low-value resistor, the control common
bus is generally at high common-mode potential
with respect to earth ground.
Some drives use non-contacting current
sensors, transformers, and optical couplers toachieve galvanic isolation of the control
circuitry. However, unless the drive is
specifically known to have an isolated analog
interface, assume it does not.

Isolation Transformers

Not all ac line-powered devices contain theinternal isolation transformers that step thevoltage down to lower operating levels neededfor electronic circuits and simultaneously protect

Measurement Computing

Power switching circuit using
SCR’s or MOSFET transistors
Internal drive control common

Bypass

DC/DC
Converter(7.10)

AC voltage
supply lines(508) 946-5100

HostcomputerDataacquisition
systemUnisolated
analogoptionComputer Digital
Analog Analog CH-HI
common common common common COMz
Tied
Low impedanceEarthgroundCurrent sense
amplifier

Motor
Shunt

Fig. 10.12. Common-mode voltage is defined as thevoltage between a signal and the localized common,
which is usually earth ground. Common-mode voltages can be quite high, depending on the signal
source. For example, a current sensing shunt in a dc
motor drive might be expected to be at line-voltage
potential, although the actual signal level is lessthan 50 mV across the shunt terminals. Measurement
techniques for small signals at high common mode
potentials call for isolation.

users from external ground faults. The power
supply common bus in transformerless devices
often connects to one side of the ac line cord. If
the system is not protected from reversing theline cord connector in the socket, the device
common line (and hence the enclosure) can beraised to a voltage level which is higher than thecommon terminal of other devices in the vicinity
or other instruments connected to the dataacquisition system. The ground fault resulting
from this arrangement effectively shorts out the
supply line and can be a hazard to operators anddestroy the equipment.
An isolation transformer lets a user safely
connect a data acquisition system input
channel, grounded through the host computer,

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CHAPTER 10 Noise Reduction and Isolation

to a low voltage signal in an ac-powered
device. The preferred isolation transformer forsuch use has a grounded electrostatic shield
between the primary and secondary windings
to minimize capacitive coupling and potentials with respect to earth ground. Thisapproach works only when the ac neutral in
the electrical system is grounded to earth.
Isolation transformers cannot interrupt thepath for the safety ground carried by theground prong of a standard 3-wire cord.
A laptop computer running on an internal
battery and not connected to other peripherals
that are tied to earth ground (such as printers)
can be a floating host safely connected to a dataacquisition system. But a better overall approachis to isolate the input signal source.
An isolation transformer usually doesn’t supply
all the isolation intended for a data acquisition
system, because the digital common in mostcomputers provides a low impedance path toearth ground as part of its ESD (electrostatic
discharge) protection scheme. But some dataacquisition devices that communicate with thehost computer through serial data links such as
RS232 and RS485 use communication isolators
specifically designed for that protocol. By
comparison, most Ethernet interfaces are transformer isolated and have ESD protection
networks on both sides of the isolation barrierreferenced to chassis, plus earth ground throughthe host computer.

Analog Isolators

The ideal solution for measuring high commonmode signals is the analog isolator. The isolator
safely measures low-level analog input signals
containing as much as 1,500 V common mode,through magnetic, optical, or capacitive devices.
The amplifiers provide both channel-to-system
isolation and channel-to-channel isolation. By
contrast, most solid-state multiplexers have nochannel-to-channel isolation beyond the standard ±10 V signal range.

Measurement Computing(508) 946-5100

The most common analog isolators are plug-in
modules. These 3-port devices require a dc power
supply and provide operating voltages for signal
conditioning and modulation circuitry on the
input side. They also provide voltage for demodulation and signal reconstruction circuits on the
output side. Most devices also provide inherent
low-pass filtering and scaling to 0 to 5 V output
levels. The wide array of available options inthese modules can simplify many complex
measurement requirements and still provide data
to the overall data acquisition system of choice,as all data acquisition manufacturers have products that accept these modules.
The isolation modules are relatively expensive
and are not likely to be used in low-cost dataacquisition systems. Low-cost systems typically don’t contain analog isolators, but many
applications require isolators in at least a few
channels. System-wise, the best place toaddress a high common-mode signal channel
is at the source for safety and signal integrity.

Wireless Techniques

Not all data acquisition systems can connect
to sensors on the test specimen with wires.
They require a form of radio communication
called telemetry. Radio transmitters andsensors are located on the device under test
and receivers are located at the data acquisition system. For example, the rotating memberof a large motor or generator can be monitored remotely and safely. The system canmonitor temperature, vibration, deflection,and speed in rpm without the type of slip
rings used in the past.
A relatively new protocol called Bluetooth isincreasingly being used for remote measurement
and control. It is a short-range wireless systemthat lets devices recognize, connect, and transfer
data among them. The devices are equippedwith special Bluetooth chips and transmit over ashort range, typically 10 m. They can transfer
data at a rate of 720 kb/sec over a frequency

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Data Acquisition Handbook

NOISE REDUCTION
Signal Averaging

Some noise reduction techniques prevent thenoise from entering the system initially, andothers remove extraneous noise from the signal.
Another technique averages several signal
samples through software. Depending on thenature of the noise and the specific averaging
method, noise can be reduced by the square rootof the number of averaged samples (RMS). But
this may require abundant samples to obtain an
acceptable measurement.
Figure(7.03)
10.13 shows
Figure 10.13
the voltage across the shorted channel when
only 16 samples of data are averaged.
Although averaging is an effective technique,it has several drawbacks. The noise present ina measurement sequence decreases as thesquare root of the number of measurements.
Therefore, in the above example, reducing the
RMS noise to a single count by averaging
alone would require 3,500 samples. As such,averaging suits only low-speed applications,and it eliminates only random noise. It does
not necessarily eliminate many other types ofannoying system noise, such as periodic noisefrom switching power supplies.

5.00E-03
4.00E-03
3.00E-03
2.00E-03
1.00E-03
0.00E+00-1.00E-03-2.00E-03-3.00E-03-4.00E-03-5.00E-03

0

0.05

0.1
0.15
Time [seconds]

Raw Signal

A filter is an analog circuit element that selectively attenuates a particular band of frequencies in an incoming signal. Filter circuits canbe passive or active. Depending on whether
the filter is low or high-pass, it determines
the frequencies that are attenuated above or
below the cutoff frequency. For example, as asignal frequency increases beyond the cutoff(508) 946-5100

Figure 10.14 (7.04)

0.2

0.25

Averaged 16 values

Fig. 10.13. Signal-averaging using software techniques virtually reduce the effective electrical noisepick up before the data acquisition system processes
the measured signal.

point of a single-pole, low-pass filter, its
attenuation increases slowly. Multiple-polefilter attenuation also increases slowly.
Multiple-pole filters provide greater attenuation beyond the cutoff frequency, but they
may introduce phase shifts that could affectsome applications. The frequency where thesignal is 3 dB down is given by the equation
shown in Figure 10.14.

Single-Pole RC FilterR

Vin

Analog Filtering

Measurement Computing

Noise Reduction: Signal Averaging

Voltsband of 2.40 to 2.48 GHz. Another system is the802.11 Ethernet-based wireless system. It often
provides physical and electrical isolation on thefactory floor and for high voltage utility linesand demolition test sites. It operates in the same
frequency range as Bluetooth and can handle
higher data transfer rates of 1 to 11 Mb/sec.

C

F-3dB =

Vout

1
2πRC

Fig. 10.14. Passive single pole filters are moderately
effective in passing a band of frequencies that contain the measured signal, while attenuating otherfrequencies containing noise.

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CHAPTER 10 Noise Reduction and Isolation

Passive vs. Active Filters

The three-pole filter shown
the example
Figurein
10.15(7.05) has
an active input with changeable configurations. The active, three-pole filter can be a
Butterworth, Bessel, or Chebyshev with cornerfrequencies up to 50 Hz. Filter properties
depend on the values of the resistors andcapacitors, which the user can change. Filters
also use switched-capacitors. This type requires
a clock signal to set the cutoff frequency. The
primary advantage of this filter is the ease of
programming the cutoff frequency.

Differential Voltage Measurement

Differential input amplifiers are most often
used in data acquisition systems because they
provide a high gain for the algebraic difference
between their two input signals or voltages,
but a low gain for the voltages common toboth inputs. Making differential voltage
measurements is another means of reducing

Comparing Single-Pole andThree-Pole Filters
0
-10
-20

Amplitude in dBV

-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
0.1

1
10
Frequency in kHz
1-pole1003-poleFig. 10.15. The three-pole filter attenuates the frequencies after the cutoff point more effectively thandoes the single-pole filter.

Noise Reduction: Low-Pass Filter

Volts

A passive filter is a circuit or device consisting
entirely of non-amplifying components, typically inductors and capacitors, which pass
one frequency band while rejecting others.
An active filter, on the other hand, is a circuitor device composed of amplifying components such as operational amplifiers, andsuitable tuning elements, typically resistors
and capacitors, which pass one frequency
band while rejecting others. Figure 10.15
compares the amplitude of a single-pole, lowpass filter with a three-pole filter. Both types
are set for a 1 kHz cutoff frequency. The
three-pole filter has a much greater attenuation for frequencies exceeding the cutoff. The
improvement in signal quality provided by
low-pass filtering is demonstrated in Figure
10.16 in which a signal containing wideband
noise passes through a three-pole filter with a
1 kHz cutoff frequency. The deviation fromthe average signal is plotted in volts. The
maximum deviation is 6 counts, and the RMS
noise is 2.1 counts.

5.00E-03
4.00E-03
3.00E-03
2.00E-03
1.00E-03
0.00E+00-1.00E-03-2.00E-03-3.00E-03-4.00E-03-5.00E-03

0

0.05

0.1
0.15
Time [seconds]

Raw data

0.2

0.25

Filtered output

Fig. 10.16. An active, three-pole low-pass filter, configured as a Butterworth, Bessel, or Chebyshev filter,
reduces the noise signal more effectively without
significant attenuation than does a passive circuit.

Figure 10.16 (7.06)
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Data Acquisition Handbook

noise in analog input signals. This techniqueis effective because often, most noise on thehigh-side input lead closely approximates thenoise on the low lead. This is called commonmode noise. Measuring the voltage difference
between the two leads eliminates this commonmode noise.

Differential-Input Amplifiers
5.00E-03
4.00E-03
3.00E-03
2.00E-03
Volts

The improvement gained with differential
voltage measurements is illustrated in Figure
10.17. It shows the same signal as Figure 10.05,
but using a differential input rather than asingle-ended input.

115

1.00E-03

0.00E+00-1.00E-03-2.00E-03-3.00E-03-4.00E-03-5.00E-03

0

0.05

0.1
0.15
Time [seconds]

0.2

0.25

Fig. 10.17. The biggest improvement in reducing
unwanted noise signals from the measured variable
comes from differential-input amplifiers. It works sowell because most of the noise on the high side duplicates the noise on the low side of the input and thealgebraic sum of the two equal parts.

Figure 10.17 (7.07)

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CHAPTER 10 Noise Reduction and Isolation

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117

Chapter 11
DIGITAL AND PULSE-TRAIN CONDITIONING

DIGITAL I/O INTERFACING
Digital Signals

Digital signals are the most common mode ofcommunications used between computersand peripherals, instruments, and other electronic equipment because they are, of course,
fundamental to the computers’ operation.
Sooner or later, all signals destined to becomputer inputs must be converted to adigital form for processing.
Digital signals moving through the system
may be a single, serial stream of pulses entering
or exiting one port, or numerous parallel lines
where each line represents one bit in a multibit word of an alphanumeric character. The
computers’ digital output lines often control
relays that switch signals or power delivered toother equipment. Similarly, digital input lines
can represent the two states of a sensor or aswitch, while a string of pulses can indicate
the instantaneous position or velocity of
another device. These inputs can come fromrelay contacts or solid-state devices.

FigureDigital
11.01 (8.01)
High Current and Voltage
I/O

Relay contacts are intended to switch voltages and currents that are higher than thecomputers’ internal output devices canhandle, but the frequency response of theircoils and moving contacts is limited torelatively slowly changing I/O signals orstates. Also, when an inductive load circuit
opens, its collapsing magnetic field generates a high voltage across the switch
contacts that must be suppressed. A diodeMeasurement Computing(508) 946-5100

across the load provides a path for thecurrent spike while the inductor’s magnetic
field is collapsing. Without the diode,arcing at the relay’s contacts can decrease
its life. (See Figure 11.01.)

Flyback Diode Protection
i
Relay
Relay
control
input

+
V2

–

Antikickback
diode

Inductive
loadiFig. 11.01. The fly-back diode clips high voltage
spikes ordinarily developed across the inductive loadwhen the control relay contacts open. Without
diodes, the high voltage arcs across the opening contacts, substantially reducing their useful life.

TTL and CMOS devices usually connect
directly to high-speed, low-level signals, suchas those used in velocity and position sensors.
But in applications where the computer energizes a relay coil, TTL or CMOS devices may
not be able to provide the needed current andvoltage. So a buffer stage is inserted between
the TTL signal and the relay coil, typically tosupply 30 V at 100 mA.

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CHAPTER 11 Digital and Pulse-Train Conditioning

An example of this type of system is an
optional card for a digital I/O instrument. It
contains an amplifier/attenuator stage,
consisting of a PNP transistor, a fly-back diode,and a resistor. (See Figure 11.02.) To energize astandard 24 V relay, an external 24 V supply isconnected to the circuit. As the internal TTL
output goes high, the transistor is biased andthe output goes low, (about 0.7 V). When the
TTL output is low, the transistor stops
conducting and the output goes to 24 V. Since
the relay coil is an inductive load, the fly-back
diode should be attached to prevent damage
during switching.

Relay Interface Card

Relay Interface Card
TTL input
User
input

Resistor in
DIP packageR
15 kΩFig. 11.03. One optional interface card comes withan adjustable attenuator to handle an input range
from 5 to 48 V in four steps. The table specifies thevoltage levels that can be selected with four different
resistors.

Flyback pull-up
voltage applied
Typical Voltage Levels & Associated Resistor Values
here Figure+V11.04A (8.04)
Flybackdiode

TTL signal

Removable
27 kΩ
pull-upresistor
200 mA
(max)

Relay
Coil
OutputPower
SourceReturnResistor Value, R
10 Ω

0 to 12 V

20 kΩ

0 to 24 V

56 kΩ

0 to 48 V

120 kΩ

Fig. 11.04. Typical voltage levels and resistor values.

DIGITAL INPUTS

L/V

Fig. 11.02. A plug-in card containing a special digital amplifier is often provided to interface a TTL
signal to a control relay. The amplifier can boost the11.04B
standard 5 V TTL signalFigure
to 24 Vdc
at 200 mA.

Figure 11.03 shows a high-voltage digital
input with an attenuator circuit. This allows
the TTL circuitry to read voltages up to 48 V.
The high-voltage signal connects to a resistive
voltage divider, which is a signal attenuator.
Selecting an appropriate resistance value R
provides a means for selecting the highvoltage level. The table in Figure 11.04 shows
the resistor values for frequently used levels.

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Inputs0 to 5 V(508) 946-5100

The methods used for interfacing digital inputs
to a computer range from simple to complex.
This section briefly discusses software-triggered,single-byte readings; hardware-paced, digital
input readings; and externally triggered, digitalinput readings.

Asynchronous Digital-Input Readings

A software-triggered, asynchronous reading isrequired when the computer periodicallysamples a digital byte or a group of bits. And
sometimes the speed and timing of the digitalinput readings are particularly critical. But thetime between readings is likely to vary whenusing the software-triggered, single-byte
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Data Acquisition Handbook

under a multitasking operating system such as
in a PC. The reason is that the time between
readings depends on the speed of the computerand other tasks that must be performed
concurrently. Variations in time between
readings can be partially compensated with
software timers, but a timing resolution of lessthan 10 ms is not guaranteed on a PC.

Synchronous Digital-Input Readings

Some systems offer hardware-paced digital
input readings. In such systems, the user sets
the frequency that the digital input port canread. For example, one system can read its
16-bit port at 100 kHz, while another operates to 1 MHz. The greatest advantage of
hardware-based digital input readings is thatthey can be implemented far more quickly
than can software-triggered readings. Finally,
devices such as these can locate the digitalinput port readings among the analog readings, providing a close correlation between
analog and digital input data.

Externally Triggered Digital-Input
Readings

Some external devices provide a digital bit, byte,
or word at a rate independent of the data acquisition system. They take readings only when
new data are available rather than at a predetermined interval. Because of this, such external
devices typically transfer data through a handshaking technique. As new digital information
becomes available, the external device issues adigital transition on a separate line, such as anExternal Data Ready or Strobe input. To interface with a device like this, the data acquisition
system must provide an input latch, which theexternal signal controls. Furthermore, a logic
signal supplied to the controlling computer
alerts it to the fact that new data are ready to bereceived from the latch.
One example of a device that operates in thisfashion has an inhibit line among its six hand-

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119

shake/control lines for notifying external
devices that the input latch is being read. This
procedure lets the external device hold new
digital information until the current read
event is performed successfully.

DIGITAL ISOLATION

Digital signals often are isolated for several
reasons: To protect each side of the system froman inadvertent over-voltage condition on the
opposite side, to facilitate communication
between devices with different grounds, and toprevent injury when circuits are attached topeople in medical applications. One common
approach for isolation is through an optical
coupler. Optical coupling consists of an LED ordiode laser to transmit the digital signal and aphotodiode or phototransistor to receive it. (SeeFigure 11.05.) Small optical couplers isolate
voltages up to 500 V. For example, this technique effectively controls and monitors digital
devices connected between dissimilar grounds.

PULSE TRAIN SIGNAL CONDITIONING

In many frequency-measurement applications, pulses are counted and compared against
a fixed time base. A pulse can be considered adigital signal because only the number ofrising or falling edges is measured. In many
instances, however, the pulse-train signal
comes from an analog source, such as amagnetic pickup.
For example, one widely used input frequency
card in a data acquisition system provides fourchannels of frequency input through two separate front-end circuits, one for true digital input
circuits and one for analog inputs. The cardconditions digital inputs of different levels, andthe analog input circuit converts a time-varying
signal into a clean digital pulse train.
Figure 11.06 shows the schematic of theanalog input, signal-conditioning path. The
front-end RC network provides ac couplingallowing all signals above about 25 Hz to pass.

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CHAPTER 11 Digital and Pulse-Train Conditioning

Optical Coupler Isolation

High

Vcc
1.8k

47k
4N28

DigitalinterfaceControl2

Address

6

Data

8

To dataacquisition
systemDigitalinputsLow
One opto-isolator

Fig. 11.05. An optical isolator decouples the digital signal source ground from the data acquisition system
ground, and protects the acquisition system inputs from unintentional high voltages.

The 11.05
selectable attenuator reduces the overall
Figuremagnitude of the waveform to desensitize thecircuit from unwanted low-level noise. Whenusing a pulse train from a relay closure, theunit provides programmable settings that let
the user select the amount of debounce timerequired. The digital circuitry monitors theconditioned pulse-train for a sustained highor low level. Without debouncing, the extra
edges in the signal produce an excessively
high and erratic frequency reading. (SeeFigure 11.07.)
Many transducers generate frequency-modulated output signals rather than amplitudemodulated. For instance, sensors that measurerotational motion and fluid flow typically fall
into this class. Photomultiplier tubes andcharged-particle detectors also are often used
for measurements that require pulse counting.
In principle, such signals could be sampledwith an ADC, but this approach generates
much more data than necessary and makes
the analysis cumbersome. Direct frequency
measurements are far more efficient.

Measurement Computing(508) 946-5100

FREQUENCY-TO-VOLTAGE
CONVERSION

Data acquisition systems measure frequency inseveral ways; they integrate a continuous wave
ac signal or pulse trains to produce a dc voltage
with a magnitude proportional to the frequency,
convert the ac voltage to a binary digital signal
with an ADC, or count digital pulses.

Analog to Pulse Converter
BNC 0.33 µF
20 kΩ

Attenuate
1:1 or 1:50

Filter100 kHz300 kHzor 30 Hz

Fig. 11.06. An interface circuit contains an RC network that significantly attenuates ac signals below
25 Hz before converting the higher frequencies to adigital pulse stream.

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Data Acquisition Handbook

Debounce Feature
Sustainedlow"Bouncy" input

Ignored
edges

a b c

Sustained
lowd e

Pulse-Train Integration
Ignorededges

Raw input
After initialsignalconditioningf

Debounced
outputTime (ms)121

Conditioned signal
has constant pulse width

Integrated
output voltage
0

10

20

30

40

50

60

Fig. 11.07. A relay contact debounce circuit filters
extraneous signals that the measurement amplifier
could mistake for rising and falling edges of a legitimate string of signal pulses.

Pulse-Train Integration
Figure 11.08

One conversion technique commonly used ina single-channel, modular signal conditioner
integrates the input pulses and produces an
output voltage proportional to the frequency.
First, a series capacitor couples the ac signal,
which removes extremely low frequency acand dc components. A comparator generates aconstant pulse width each time the input
signal passes through zero. The pulse thenpasses through an integrating circuit such as a
low-pass filter and generates a slowly changing
signal level at its output, proportional to the
input frequency. (See Figure 11.08.)
The response time of the frequency-to-voltage
converter is low – the inverse of the cutoff
frequency of the low-pass filter. This cutoff
frequency should be much lower than the
input frequencies being measured, but highenough to provide the required response time.
As the measured frequency approaches thecutoff frequency however, significant ripple in
the output becomes a problem as shown in
Figure 11.09.
An external capacitor selects the time constantfor an IC dedicated to frequency to voltage con-

Measurement Computing(508) 946-5100

Fig. 11.08. In one type of frequency-to-voltage converter, an integrator circuit converts a string of constant-width pulses of varying repetition rate into asmoothly changing analog voltage. The analog signal is proportional to the input frequency.

version. The circuit can measure signals invastly different fixed frequency ranges, but thecapacitor must be changed to change a frequency range. Unfortunately, such frequencyto-voltage converters work relatively poorly forfrequencies below 100 Hz because a low-passfilter with a cutoff frequency under 10 Hz
requires an excessively large capacitor.

Digital-Pulse Counting

Another type of conversion techniquemeasures the frequency of a string of digitalpulses or an ac-coupled analog signal voltage.
It outputs a dc voltage level proportional to

Output Ripple
Low frequency
pulsesIntegrated
outputFig. 11.09. The pulse-train integrator handles only
frequencies above a certain cutoff, due to the integrator’s time constant. Frequencies below cutoff produce
output ripple if allowed to pass.

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122

CHAPTER 11 Digital and Pulse-Train Conditioning

the input frequency, similar to the integrator
described above. However, the dc level herecomes from a DAC output. Front-end circuitry
converts the incoming analog or digital signal
into a clean pulse train, devoid of relay contact
bounce, high-frequency noise, and otherunwanted artifacts before it reaches the DAC.
(See Figure 11.10.)
For example, the analog input channel of atypical frequency-input data acquisition card
contains a low-pass filter with a selectable
cutoff frequency of 100 kHz, 300 Hz, or 30 Hz.
It measures frequencies from 1 Hz to 100 kHz
for signals ranging from 50 mV to 80 Vp-p.
The digital input circuit measures ±15 Vdc
signals from 0.001 Hz to 950 kHz, dc-coupled
to a TTL Schmidt trigger circuit. The cards
typically come with pull-up resistors for use
with relays or switches.
A microcontroller accurately measures a total
period consisting of several cycles extending
over one user-selectable minimal period, which
determines the frequency resolution. The micro-

controller computes the frequency from themeasured period and converts it to a command
for a DAC, which, in turn, provides the dc levelto the data acquisition system. The dc output of
the DAC drives the input of an ordinary dc
signal conditioner, and the software converts
the dc level to an equivalent frequency reading.
This method allows extremely low frequencies
to be measured over an exceptionally wide
range, and the output update can be relatively
fast. Moreover, the frequency range can be
programmed, letting the expected frequencies
use the entire ADC range.
The output range of the DAC is +5 to -5 V. The
minimum frequency selected by the user
becomes the -5 V output while the maximum
frequency becomes +5 V. Virtually any frequency
bandwidth may be selected such as 500 Hz Fmin
to 10 kHz Fmax, or 59.5 Hz Fmin to 60.5 Hz
Fmax. With an ADC of 12-bit resolution, thelower bandwidths will have higher resolution
than higher bandwidths, simply because therange from -5 to +5 Vdc is partitioned into 4096

Digital Pulse Counting
Jumpers shown configured
for analog input mode

Schmitt-trigger
bufferDigital input mode
BNC
Frequency
measurement

Analog input mode
Selectableattenuatora

DAC

To dataacquisition
systemLPF
fMicroprocessor-basedfrequency measurement circuit

Low pass
filterChannel 0

Fig. 11.10. In the other type of frequency-to-voltage converter, the input card contains an input coupling capacitor to filter low frequencies, an attenuator, a low pass filter, and a microprocessor to calculate the frequency. The
signal is then connected to a DAC before feeding into the data acquisition system.

Measurement Computing(508) 946-5100

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Data Acquisition Handbook

parts regardless of bandwidth. The one Hz
bandwidth is divided into 4096 parts, whichyields a resolution of 1/4096 Hz or about
0.00244 Hz. For the 100 kHz bandwidth, theresolution becomes 24.41 Hz.
The resolution is 12 bits over all ranges, but the
update time depends on the range selected.
From 1 Hz to the user-defined maximum upper
range boundary, the voltage conversion updateis 2 to 4 ms or the period of the input frequency,
whichever is greater. For a range of 0 to 10 kHz,the update rate is 2 to 4 ms, and for a range of
0 to 60 Hz, the output updates every cycle or16.6 ms. As the conversion range becomesnarrower, from 49 to 51 Hz, for example, thetime to resolve the 2 Hz differential to 12-bit
resolution increases. In this case, the conversion time is approximately 59 ms.
In addition to the low-pass filter, a predefinedhysteresis level is built in to help prevent false
counting caused by high-frequency noise. A
debounce time can be programmed from
0.6 ms to 10 ms for handling electromechanical devices such as switch or relay contacts
that bounce or chatter while switching.

Frequency Measurement byGated-Pulse Counting

Gated-pulse counting can measure frequencies
much more accurately than frequency-tovoltage conversion methods. Gated-pulse
counting methods count the pulses thatappear over a specified period of time. Dividing
the number of pulses by the counting interval
determines the frequency, and the error can be
as low as the inverse of the counting interval.
For example, if the counting interval is two
seconds, the error can be as low as 0.5 Hz.
Many data acquisition systems include TTLcompatible counter/timer ICs that can performgated-pulse, digital-level input, however, they
are unsuitable for unconditioned analog signals.
Fortunately, many frequency-output devices

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123

feature a TTL output option. Some products use
a counter/timer IC, which contains five counter/timers. Many counter/timer ICs generally use
an oscillator built into the data acquisition
system, or an external oscillator. Such ICs
usually have several channels available to assist
counting applications. Each channel contains
an input, a gate, and an output. The simplest
counting method only uses the input, and the
PC is programmed to periodically read andreset the counter. The weakness in this approachis the uncertainty in the timing interval.
Variations crop up in the execution speeds ofthe functions that begin and end counting. In
addition, the function call that delays executionof the program for 50 ms runs under aninaccurate software timer. These two effects canrender a short-counting interval, frequency
measurement useless. However, the technique
is usually sufficient for counting intervals
greater than one second.
Gating can attain greater accuracy because thegate controls the counting interval.
Consequently, frequency measurements are
independent of any software timing issues.
The gate can be configured so that pulses arecounted only when a high-level signal enters
it. Similarly, the gate can start counting when
it detects one pulse and stop counting when it
detects another.
A disadvantage of gated-pulse counting is thatit requires an extra counter to provide thegate. However, in multiple channel applications, a single counter can provide the gate formany channels. For example, in a five-channel
system, four channels count while one channel
provides a gate.

Timing Applications

A counter/timer also can be used in a dataacquisition system for timing applications. A
clock signal connected to the input of achannel and using the input signal as a gateworks well. The method requires that the gate

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CHAPTER 11 Digital and Pulse-Train Conditioning

be configured for counting when the gate
input is high. A similar technique can measure
the length of time elapsed between two pulses
by configuring the gate to begin counting at
the first pulse and end counting at the second.

Measurement Computing(508) 946-5100

Because a 16-bit counter overflows at 65,535
counts, the maximum pulse width measurablewith a 1 MHz clock is 65,535 ms; a longer pulse
overflows the counter. However, a clock slower
than 1 MHz can be used for longer pulses.

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125

Chapter 12
TEDS: TRANSDUCER ELECTRONIC DATA SHEETSSMART SENSORS

Certain types of transducers are classified as
smart sensors. They contain a Smart Transducer
Interface Module (STIM), which in turn,
contains an onboard EEPROM memory IC,
called TEDS, Transducer Electronic Data Sheet.
(See Figure 12.01.) The TEDS electronicallystores information regarding the transducer’s
characteristics and parameters such as type ofdevice, manufacturer, model number, serial
number, calibration date, sensitivity, reference
frequency, and other data. The TEDS also
stores the calibration coefficients andfrequency response for a transducer in termsof a table or an algorithm.

TEDS: Principal Parts
TII
Network
Transducer
Independent
I/F

STIM
XDCR

ADC

XDCR

DAC

XDCR

DI/O

XDCR

?

Address
logic

TEDS(part 2)

NCAP
(part 1)

Network-Capable
Application Processor

Transducer Electronic Data Sheet

Fig. 12.01. Three principal parts comprise smart
sensor systems: the Smart Transducer Interface
Module STIM, the Transducer Electronic Data Sheet
TEDS, and the Network-Capable Application
Processor NCAP.

Measurement Computing(508) 946-5100

Not all sensor manufacturers provide this feature,
but those that do are encouraged to follow an
evolving standard, IEEE 1451.0. One part of thestandard, 1451.4, defines the TEDS format,
channel identification formats, electrical interface, and read and write functions for accessing
the TEDS and transducer data. The specification
also defines the data set, that is, the number ofsamples acquired for one command that varies
from 0 to 65,535 samples per set.
The standard does not specify requirements
for signal conditioning, signal conversion, or
how applications can use the TEDS data.
However, the signal conditioners and otherinterface hardware used with these smart
sensors must provide for an option or include
circuits for TEDS communications under the
direction of a software module specificallyintended for this purpose. Another special
module that works with TEDS is usually part ofthe data acquisition system’s application software package that collects, stores, and displays
the measured variables. The hardware automatically scales the readings and sets therange according to the data stored in thesensor. This is typically done for each TEDSassociated channel.
TEDS capability was originally intended for
piezoelectric sensors such as accelerometersand pressure sensors, but it now includes allcommon analog sensors and actuators, such as
MEMS (micro-electromechanical sensors),
accelerometers, pressure transducers, andtemperature sensors with two-wire and mixedmode (analog and digital) input/output.

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CHAPTER 12 TEDS: Transducer Electronic Data Sheets

For two-wire analog sensors and actuators(called Class 1), the output signal is generally
coupled to the signal conditioner or driver
while the TEDS data are enabled and read out
with a dc bias voltage applied to the same two
wires. (See Figure 12.02.) The transducers with
mixed-mode capability (called Class 2) also
communicate digitally with the TEDS memory.
(See Figure 12.03.) The analog part is the signal
representing the measured variable, and thedigital interface communicates with theembedded EEPROM. The TEDS architecture
contains standard templates for common
transducers, and custom sub-templates that
manufacturers can use for defining special
parameters and custom requirements.

Sensor+Current
sourceFigure 12.03 Analog
signalA

TEDS EEPROM

Single-wire systemsmart sensor

Digitalsignal
-5 V
Data acquisition
systemFig. 12.02. Class 1 sensors have only two leads thatconnect to the input of the STIM. The analog signal
voltage and any required biasing voltage share thesame pair of leads.

The TEDS file may be contained onboard thesensor in the EEPROM, or off board in areserved file in the data acquisition system. A
number of transducer manufacturers plan tolaunch an Internet-accessible site where theylist sensors that presently do not contain a
TEDS memory IC. Users may enter the serial
number of their sensor and obtain an equivalent

Measurement Computing

Class 2 interfaceplug-and-play sensor
AnalogExcitation
_
+

V

A

TEDS EEPROM

DigitalsignalDigitalSmart sensor
multi-wire system

AnalogsignalData acquisition
systemWheatstone Bridge strain gage

Class 1 Sensors: Analog
Class 1 interface
Plug-and-play sensor

Class 2 Sensors: Mixed Mode(508) 946-5100

Fig. 12.03. Class 2 sensors use multiple leads toconnect between the sensor and STIM. In the case ofa Wheatstone Bridge circuit, for example, four leads
connect the sensor and the other pair connects to the
onboard TEDS EEPROM, which illustrates themixed-mode application.

TEDS specification to download. This lets
application software programs work with botholder sensors and newer TEDS-equipped sensors.

NETWORKSThe original intent of the TEDS concept is tohandle arrays of smart sensors and actuators
over various types of networks. Different typesof control networks now exist for both analog
and digital communications, but most are
intrinsically incompatible with one another.
And not all transducer manufacturers areeither able or willing to provide a uniquesensor for each kind of network bus. As aconsequence, they are increasingly turning todigital networks controlled with microprocessors to develop a universally accepted interface
that is more economical for them to interconnect systems, networks, and instruments. The

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Data Acquisition Handbook

guiding document is the evolving IEEE 1451
standard, which also allows the new devices tobe compatible with older and existing systems.

SYSTEM ARCHITECTURE

The IEEE has partitioned the P1451 standard
into five major parts. Each part addresses a
different facet or interface of the project thatenables sensor manufacturers to design andbuild new sensors compatible with all
networks. Each part is defined as follows:Part 1. Network-Capable Application
Processor (NCAP) Information Model

The purpose of this part is to define an object
model for smart transducers that connect to a
network, and specify a software interface towork with the transducer’s components. The
major components of this model include the
NCAP block, function block, and transducer
block. The NCAP processor connects the
network to the transducer modules, and eachdifferent network requires a different NCAP
physical interface. The smart transducer object
model interfaces to the NCAP processor and to
the transducer block. The interface to thetransducer block contains details of the transducer hardware in a program model, and theinterface to the NCAP block contains details
describing the network protocols.
The NCAP’s primary purpose is to communicate between the STIM and a particular
network. The NCAP also computes calibration
corrections and converts between values inmetric units and values coming from the
STIM’s DACs or going to the ADCs.

Part 2. Transducer to Microprocessor
Communications Protocols and Transducer
Electronic Data Sheet (TEDS) Format

Part 2 defines the details of a TEDS, including
the format for storing data in a small (256-bit to4 kb) EEPROM and the interface between the
NCAP processor and the transducers. The TEDS
contains the information needed by the

Measurement Computing(508) 946-5100

127

software to convert the sensor values such as
volts or resistance to physical units, such as
force in pounds or acceleration in gs. TEDS is
also part of the Smart Transducer Interface
Module. STIM includes the ADCs, DACs, digital
I/O, and triggers that connect to the transducers.

Part 3. Digital Communications and TEDS
Formats for Distributed Multidrop Systems

This part is intended to define the TEDS
format and the standard for the interface
between multiple transducers in a multidrop
network. Some of the issues dealing with amultidrop system include automatically identifying the transducer when it connects to thebus, and how quickly the system recovers after
a short power dropout.

Part 4. Mixed-Mode Communication
Protocols and TEDS Formats

Mixed-mode communications deals with theissue of using two-wire I/O, which shares bothsignal and digital interfaces vs. a multi-wiresystem where signal and digital communications are handled on separate wires or ports.
Small analog sensors contain TEDS that let thesensors interface on the network through thesame two wires they use for signals. The digital
TEDS data uses the same two wires.

Part 5. Wireless Communication Protocols
and TEDS Formats

At the time of this writing, the IEEE P1451
standards committee study group had defined
the details of the interfaces for wirelesscommunications, but not the modules. The
intent of the standard is to separate the physical layers from the upper layers of the protocol
stack. Significant progress on the specification
is expected in 2004.

TEDS COMPOSITION

The data structure for the TEDS works wellwith a variety of sensor types. Regardless ofthe sensor’s principle of operation, the structure contains three major subdivisions;

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128

CHAPTER 12 TEDS: Transducer Electronic Data Sheets

TEDS Table
Meta-TEDS
Provides operational parameters for the transducer
Transducer-channel TEDS
Provides operational parameters for the transducer
Calibration TEDS
Provides calibration coefficients for the transducer
Text-based TEDS
Provides a description of the device for the operator, human readable in XML
Frequency-response TEDS
Provides frequency response for a transducer and associated electronics in a table
Transfer-function TEDS
Provides frequency response for a transducer and associated electronics in algorithm form
End-user, application-specific TEDS User writeable block of non-volatile memory. Contents defined by user.
Commissioning TEDS
User writeable block of non-volatile memory. Intended to provide an identifier for the transducer.
Manufacturer-defined TEDS
Format defined by the manufacturer
Physical TEDS
Defined for the particular physical layer, not included in P1451.0. How to access it is defined byIEEE P1451.0.

Fig. 12.04. Table 1. Basic TEDS Table.

the Basic TEDS; the Standard TEDS (subtemplate), and the User Area. (See Figure 12.04,
Tables 1, 2, and 3.)

The Basic TEDS contains the manufacturer’s
identification, model identification, version
letter, and the serial number, which are common
to all types of sensors. The Standard TEDS,
Figure 12.04A
however, contains data unique to the sensor’s
principle of operation. For instance, a piezoelectric accelerometer differs from a strain gageload cell, so the Standard TEDS EEPROM
contains some common and some uniqueinformation. Both sensor TEDS list Calibration

TEDS Configuration: Accelerometer
Basic TEDSManufacturer ID
Model ID
Version Letter
Serial Number

25
5226
A
11521S

Standard TEDS
(Subtemplate)

Calibration Date
Sensitivity @ Reference
Reference Frequency
Reference Temperature
Measurement Range
Electrical Output
Quality Factor
Temperature Coefficient
Direction (X, Y, Z)

Jan. 6, 2004
1.12E+03 mV/g
110.0 Hz
25˚C
±50 g
±5 V
-0.27%/˚C
X

Sensor Location
Calibration Due Date

Matrix A: B-3
Jun. 15, 2004

User Area

Fig. 12.04. Table 2. Example for an Accelerometer.

Figure 12.04C

Measurement Computing(508) 946-5100

Date, Measurement Range, and Electrical
Output, but they differ in as many as 12 otherparameters. The accelerometer has 6 parameters that the strain gage lacks, and the strain
gage has 6 parameters the accelerometer lacks.
The User Area contains information such as
Sensor Location, Calibration Due Date, andCalibration Table.

TEDS Configuration: Strain-Gage Load Cell
Basic TEDSManufacturer ID
Model ID
Version Letter
Serial Number

56
350A
E
150-00878TStandard TEDS

Calibration Date
Measurement Range
Electrical Output
Bridge Impedance
Measurement Range
Excitation, Nominal
Excitation, Minimum
Excitation, Maximum
Response TimeMar. 15, 2004
±250 lb±5 mV/V500 Ω±75 g
24 Vdc10 VdcCalibration TEDS Calibration Table

User Area-2.06 mV/V, -150 lb-1.02 mV/V, -75 lb
0.03 mV/V, 0 lb
0.95 mV/V, 75 lb
1.89 mV/V, 150 lb

Sensor Location
L23-3
Calibration Due Date Jun. 15, 2004

Fig. 12.04. Table 3. Example for a Strain-Gage
Load Cell.

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129

Index

A

Absolute Encoders 99
AC Voltage 24
Active Filters 49, 114
ADCs
Accuracy and Resolution 6
Accuracy vs. Resolution 6
Accuracy vs. System Accuracy 8
Integrating Dual Slope 4
Output Averaging 11
Sample-and-Hold 16
Sigma-Delta 5
Successive-Approximation 3
Types 3
Voltage-to-Frequency 4
Aliasing 17
Amplification 39
Amplifiers
Charge 90
Common-Mode 44
Differential 42
Instrumentation 44
Integrated Instrumentation 47
Inverting 41
Isolation 53
Non-Inverting 42
Operational 40
Programmable Gain 43
Programmable Gain Instrumentation 47
AnalogDigital Conversion 3
Filtering 113
Isolation Modules 54
Isolators 112
Anderson Loop 32
Asynchronous Digital Input Readings 118
Attenuation 51
Average Voltage 25

Measurement Computing(508) 946-5100

B

Balanced Differential Dividers 52
Basics of RTDs 73
Basics of Thermistors 78
BridgesCircuits 82, 83
Full 82
Half 83
Quarter 83
Wheatstone 32
Buffered Voltage Dividers 52

C

Calibration 87
Charge Amplification 90
Circuit Protection 108
Cold Junction Compensation 66
Common-Mode
Amplifiers 44
Limitations 27
Voltage Measurements 34
Voltage Rejection 108
Compensated Voltage Dividers and Probes 52
Constant Current/Voltage Drop 31
CrosstalkDynamic 106
Static 106
Current Transformers 28
Current/Measurements 26

D

Data Acquisition
Front Ends 39
PC-Based 2
Systems 1, 108
DC Voltage 23
Differential
Amplifiers 42
Measurements 34, 114

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130

INDEX

DigitalCommunications and Format 127
I/O Interfacing 117
Inputs 118
Isolation 119
Methods of Isolation 56
Pulse Counting 121
Signal Conditioning 117
Signals 117
Displacement and Position Sensing 95
Displacement, Linear 95
Dividers
Balanced Differential 52
Buffered Voltage 52
Compensated Voltage 52
Methods 31
Dynamic Crosstalk 106

E

Effective Number of Bits 9
Effective Voltage 25
Electrical Measurements 23
Encoders
Absolute 99
Errors 100
Incremental 99
Linear 98
Quadrature 100
ENOB Test 10
ESD Protection 64
Externally Triggered Digital Input Readings 119

F

Filters
Active 47, 49, 114
Analog 113
High-Pass 49
Low-Pass 48
Passive, 47, 49, 114
Switched-Capacitor 50
Floating Data Acquisition Systems 108
Fourier Transforms 17
Four-Wire RTDs 74, 75, 76, 77
Frequency Measurement 123
Frequency-to-Voltage Conversion 120
Full-Bridge Circuit 82
Fundamental Signal Conditioning 39

Measurement Computing(508) 946-5100

G

Gated-Pulse Counting 123
Gradient Nature of Thermocouples 65
Ground Loop
Creation 105
Elimination 105
Symptoms 105
Loops 104
Grounding
Conflicts 103
Robust Instrumentation 104
Safety 103

H

Half-Bridge Circuit 83
Hall-Effect Sensors 29
Hardware Linearization 62
Hazards to Instrumentation Circuits 63
High
Common-Mode Amplifiers 44
Common-Mode Voltage Measurements 34
Common-Mode Voltage Rejection 108
Current and Voltage Digital I/O 117
DC Voltage 23
Pass Filters 49
Voltage 24
Voltage Dividers 52

I

Incremental Encoders 99
Inherently Isolated Sensors 57
Insertion Loss 27
Instrumentation Amplifiers 44
Instrumentation-Level DC Voltage 23
Integrated Instrumentation Amplifiers 47
Integrating ADCs: Dual Slope 4
Introduction to Data Acquisition and
Signal Conditioning 1
Inverting Amplifier Stages 41
Isolated Data Acquisition Systems 108
Isolation
Amplifiers 53
Analog Modules 54, 112
Basics 53
Digital 119
Methods 56
Transformers 111

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Data Acquisition Handbook

K

Non-Inverting Amplifier Stages 42
Nyquist Theorem 17

LO

Kelvin Connections, Examples 37
Linear
Displacement 95
Encoders 98
Variable Differential Transformers 96
Linearization
Hardware 62
Software 61
Thermistors 78
Load Cell Calibration 87
Low
AC Voltage 24
DC Voltage 23
Low-Impedance Transducers 91
Low-Pass Filters 48

MMeasurements
Current 26
Differential, Voltage 33, 34, 114
Digital Input 118
Electrical 23
Frequency 123
High Common-Mode Voltage 34
Resistance 30
RTD 73, 74
Single-Ended 34
Strain 81
Temperature 65
Thermistor 78, 80
Thermocouple 70
Valid 23
Mixed-Mode Communications Protocol 127
Multiplexing
Fundamental Concepts 13, 15
Sample Rates 15
Source Impedance 16
Speed 14N

NCAP Information Model 127
Networks 126
Noise
Control 103
Environments 70
Histograms 9
Reduction and Isolation 103

Measurement Computing(508) 946-5100

131

Operational Amplifiers 40
Outlet-Level AC Voltage 24
Overload Protection 64

P

Passive Filters 49, 114
Passive vs. Active Filters 49, 114
PC-Based Data-Acquisition Equipment 2
Peak Voltage 25
Piezoelectric Transducers 90
Position, Rotational 98
Power Transducer Interfacing 29
Programmable Gain Amplifiers 43
Programmable Gain Instrumentation
Amplifiers 47
Pulse-Train Integration 120
Pulse-Train Signal Conditioning 117, 119

Q

Quadrature Encoders 100
Quarter-Bridge Circuit 83

R

Relays 13
Resistance
Basic Measurements 30
Small Deviations 31
RMS Voltage 25
Rotational Position 98
RTDs
Measurement Approaches 73, 74
Basics 73
Two-Wire 74
Three-Wire 74, 75, 78
Four-wire 74, 75, 76, 77

S

Safety Concerns 23
Sample-and-Hold ADCs 16
Sampled-Data Systems 13
Sensors
Displacement 95
Hall-Effect 29
Inherently Isolated 57
Low Impedance 91
Piezoelectric 90
Position 95
Smart 125

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132

INDEX

Sensors (cont’d)String Potentiometer 95
Ultrasonic 97
Vibration/Sound 89
Sequence vs. Software-Selectable Ranges 14
Settling Time 21
Shield Installation 107
Shielded Wiring 106
Shunts 27
Sigma-Delta ADCs 5
SignalAveraging 113
Conditioning 1, 39, 117
Pulse Trains 119
Sensors and Signal Conditioning 1
Single-ended and Differential Measurements 34
Single-Ended Measurements 34
Small Resistance Deviations 31
Smart Sensors 125
Software Linearization 61
Solid-State Switches/Relays 13
Static Crosstalk 106
StrainBasics 81
Measurement/Configurations 81
Strain-Diaphragm Pressure Gages 87
Strain-Gage
Accelerometers 89
Calibration Approaches 87
String Potentiometer Transducers 95
Successive Approximation ADCs 3
Switched-Capacitor Filters 50
Synchronous Digital Input Readings 119

T

TEDS Composition 127
TEDS System Architecture 127
Temperature Measurements 65
Thermistor
Basics 78
Linearization 78
Measurement Approaches 78, 80
Thermocouple
Basics 65
Cold-Junction Compensation 66
Concerns 71
Gradients 65
Measurement Pitfalls 70
Type Mixing 68

Measurement Computing(508) 946-5100

Three-Wire RTDs 74, 75, 78
Timing Applications 123
Transducers
Hall-Effect 29
Low-Impedance 91
Piezoelectric 90
Power 29
String Potentiometer 95
125Transducer Electronic Data Sheets (TEDS)Transducer to Microprocessor Protocol 127
Transformers
Current 28
Isolation 111
Linear Variable Differential 96
Two-Wire RTDs 74

U

Ultrasonic Sensors 97

V

Valid Measurements 23
Vibration & Sound Sensors 89
Vibration and Sound 89
Voltage
AC 24
Average 25
Common-Mode Measurements 34
Common-Mode Rejection 108
Conversion 4, 120
DC 23
Differential Measurements 114
Dividers, Compensated 31, 51, 52
Drop 31
Effective 25
High 23, 24, 52, 117
Instrumentation-Level 23
Low AC and DC 24
Outlet-Level AC 24
Peak 25
RMS 25
Voltage-to-Frequency ADCs 4

W

Wheatstone Bridges 32
Wireless Communication Protocol 127
Wireless Techniques 112

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Data Acquisition Handbook

133

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