Regular Papers

Compact Six-sector Antenna Using Three Intersecting Dualbeam Microstrip Yagi-Uda Arrays with Common Director
Naoki Honma†, Tomohiro Seki, Kenjiro Nishikawa,and Shuji Kubota
Abstract
This paper presents a novel compact planar six-sector antenna suitable for wireless terminals. Our new
design yields low-profile and extremely compact multisector antennas because it allows microstrip YagiUda array antennas to share director elements. Two microstrip Yagi-Uda array antennas are combined toform a unit linear array that has a feed element at each end: only one of them is excited at any one time
and the other is terminated. A numerical analysis shows that applying a resistive load to the feed port of
the terminated element is effective in improving the front-to-back ratio. We investigated the radiation
pattern of a six-sector antenna, which uses three intersecting unit linear arrays, and identified the optimum termination condition. From the results of measurements at 5 GHz, we verified the basic properties of our antenna. It achieved high gain of at least 10 dBi, even though undesired radiation was suppressed. A radiation pattern suitable for a six-sector antenna could be obtained even with a substrate having a diameter of 1.83 wavelengths, i.e., an area 75% smaller than that of an ordinary antenna with six
individual single-beam microstrip Yagi-Uda arrays in a radial configuration. We also fabricated an antenna for the 25-GHz band and found that it has potential for high antenna performance, such as antenna
gain of over 10 dBi even in the quasi-millimeter band.

1. Introduction
Studies on high-speed indoor wireless access systems, which use the microwave or millimeter bands
are being actively pursued [1], [2]. These systems
must reduce the effect of multipath fading in order tomaintain high data transmission speeds. Narrowbeam antennas and adaptive arrays can overcome thisproblem, and the effects of these antennas in wirelesscommunication systems have been studied [3]–[6]. A
narrow-beam antenna is the easiest way to improvethe data transmission speed because a fixed-beamantenna can be used and it achieves a relatively highgain. A multisector antenna comprising several narrow-beam antennas with different beam directions is† NTT Network Innovation Laboratories
Yokosuka-shi, 239-0847 Japan
Email: honma.naoki@lab.ntt.co.jp

72

one way to provide wireless connections with a highdata transmission speed because the terminals oraccess points can switch the beam direction so as todirect the beam to the desired signal [7]–[11].
Mobile terminals usually utilize only one sector ofa multisector antenna at any one moment because
only one connection to one access point is necessary.
This requires the use of a sector switching mechanism. A sector antenna with too few sectors provides
only a relatively low gain enhancement effect, which
may be canceled by the loss that occurs in the sector
switching mechanism in the feed circuit. On the other
hand, a sector antenna with many sectors requires acomplicated multiport sector switching mechanism.
Therefore, terminals are likely to use four to six sectors [7], [10], [11]. Moreover, multisector antennas
for mobile terminals must be small and low-profile
because of the recent downsizing of mobile terminals
for enhanced portability.
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A low profile multisector antenna can be fabricatedusing radially arranged microstrip Yagi-Uda arrays[12]. A microstrip Yagi-Uda array comprises an
exciting microstrip antenna and several parasitic
microstrip antennas arranged on the same substratesurface. It offers high gain [13]. Furthermore, thisantenna can be fabricated using etching techniques,
which enables mass production. However, it usuallyrequires a large substrate area because each sector
requires an individual array antenna. The other problem with the microstrip Yagi-Uda array is the difficulty in obtaining a sufficient front-to-back ratio(F/B). The high level of undesired radiation degrades
the data transmission speed [5].
In this paper, we describe a new compact planar
six-sector antenna based on a microstrip Yagi-Uda
array with common director elements that is suitable
for mobile terminals. One of the key ideas for reducing the antenna size is a dual-beam microstrip YagiUda array—a new invention for the terminal sector
antenna. In this antenna, the parasitic elements areshared between two opposite-facing microstrip YagiUda arrays, enabling the size of the array antenna to
be reduced to almost half that of two individual
microstrip Yagi-Uda arrays. Another key idea is further element sharing among the multiple linear YagiUda arrays because three dual-beam Yagi-Uda arrays
are necessary to obtain six beams. That is, the parasitic elements are to be shared not only by two beams,
but also by all six beams. This is made possible byplacing a hexagonal parasitic element at the center ofthe linear array, where the three unit linear arrays
intersect. The optimized termination condition at thefeed ports of the quiescent exciting elements is used
to improve the effect of the coupling among the intersecting linear arrays. These techniques of element
sharing and feed port termination yield an extremelycompact six-sector antenna.
Section 2 briefly describes the basic idea of thedual-beam microstrip Yagi-Uda array antenna, inwhich the two exciting elements share parasitic elements. Section 3 describes the geometry and mechanism of the six-sector antenna, in which the parasitic
elements are shared among six beams. Section 4shows a numerical analysis of the six-sector antenna.
We investigated the impact of the intersecting linear
arrays on the radiation pattern of a six-sector antenna
under various termination conditions. Section 5
describes experiments conducted on a fabricated sixsector antenna operating at 5 GHz to confirm thevalidity of the design. Section 6 describes the resultsof a trial in which this antenna was applied to the 25Vol. 5 No. 3 Mar. 2007

GHz band. The 25-GHz band is being considered foradvanced wireless local area networks (WLANs)
because more frequency channels will be needed inorder to offer high-speed data transmission to many
more users [14].
2. Dual-beam microstrip Yagi-Uda Array with
common parasitic elements
2.1 Basic geometry and mechanism
First of all, we present the idea and basic propertiesof the dual-beam Yagi-Uda array, which is the unit
component of the six-sector antenna. We mainly
examined the influence of the termination conditionof the quiescent element and the number of parasitic
elements.
The geometries of a conventional microstrip YagiUda array and a dual-beam microstrip Yagi-Uda
array, are shown in Figs. 1(a) and (b), respectively.
The microstrip antenna elements are formed on adielectric substrate, which has ground metallizationon the rear side. These elements are rectangular
patches. Each array antenna has m parasitic elementsaligned on the x-axis. The exciting element with thefeed port at the end of the array is slightly larger thanthe parasitic elements.
In the figure, a and a′ are the lengths of the long and

Wx
g
a

y

g

φ

d

Wy a ′

d′

···

m

Port

λ0/4

x

z

(a)

Wx
g

g
a

φ

d′

d

Wy a ′
λ0/4

y

g
···

Port #1

Port #2

m

λ0/4

z

h
···

εr , tanδ

x

z

y

θ

x

(b)

Fig. 1. Unit linear array configurations: (a) Top view ofconventional microstrip Yagi-Uda array and (b) top
and side views of dual-beam microstrip Yagi-Uda
array.

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short sides of the exciting element, respectively, d andd′ are the lengths of the long and short sides of eachparasitic element, respectively, g is the gap between
elements, h is the thickness of the substrate, Wx and
Wy are the lengths of the long and short sides of thesubstrate, and εr and tanδ are defined as the relative
dielectric constant and the loss tangent of the substrate, respectively.
The dimensions and material properties are set to a
= 0.34 λ0, a′/a = 0.6, d′/d = 0.5, d/a = 0.92, g = 0.005
λ0, h = 0.027λ0, εr = 2.2, and tanδ = 0.0008, where λ0
is the wavelength in vacuum. The length of the shortside of the rectangular substrate is set to Wy = λ0/2,
and Wx, which is the length of the long side, is set sothat the distance between the edge of the substrate
and the center of the end element in the linear array is
λ0/4.
The dual-beam microstrip Yagi-Uda array has two
exciting elements, one at each end of the linear array.
Each exciting element has a feed port, but only one
port is driven at any one time. That is, when one port
is excited, the other port is terminated.
In these antennas, the beam extends away from theexciting element over the parasitic elements, which
work as a director. Therefore, the antenna shown in
Fig. 1(a) can work as a single-beam Yagi-Uda array,and the antenna shown in Fig. 1(b) can work as adual-beam Yagi-Uda array if the feed ports are
switched.
2.2 Numerical analysis
The impact on antenna performance of the port termination condition and the number of parasitic elements was numerically analyzed by the method of

moments (MoM). In this numerical analysis, only thedielectric layer of the substrate was assumed to be an
infinite plane to simplify the analysis. The positionsof the feed ports were optimized to minimize thereflection of the incident power in both models. The
impedance of the terminated port #2, ZL (load impedance), was set to the characteristic impedance of thefeed line Z 0; i.e., ZL = Z 0 = 50 Ω.
To evaluate the radiation performance as a planar
sector antenna, we defined F/B and the conical radiation pattern. In this paper, F/B is defined as the ratioof the directivity of the direction of maximum radiation to that of the direction of the maximum lobe in
the range of ±60° from the opposite direction. However, since the directions of both the mainlobe and thebacklobe are tilted above the ground plane [12]–[13],F/B obtained in the usual way is underestimated. For
this reason, we created a new definition of F/B, suitable for the configuration in Fig. 1. Our definition of
F/B is shown in Fig. 2. The two cones are symmetrical and share the same apex. The mainlobe lies inside
one cone. F/B is defined as the ratio of the maximum
level in one cone to that in the direction within theother cone.
The definition of the conical plane radiation pattern
is shown in Fig. 3. We assume a conical plane including the direction of the peak radiation. The conicalplane radiation pattern can be defined by the patternobtained by cutting the three-dimensional patternwith an ideal conical plane. By using the conicalplane radiation pattern, we can define the conicalplane beamwidth as the range of the angle that limits
the gain decrease from the maximum gain to within 3dB.
The radiation patterns were calculated while varyz θ

θ°

y
A

Cross-section pattern with conical plane(Conical-plane pattern)Bφx
120˚°˚
120

Peak direction

F/B
120˚°˚
120
Radiation patternzyConical plane
Peak gainθ(Peak gain) – 3 dB

Fig. 2. Definition of F/B in this paper.

74

Conical-plane beamwidthxFig. 3. Definition of conical-plane beamwidth.

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|E θ | (dB)

Open
0
–5
–10
–15
–20
–25
–30

θ°

0

330

30
60
Short
90

240
Z L = Z0

120
210

150

Beamwidth in conical plane (degrees)

Regular Papers

90
Conventional antenna

80
70
60
50
40

Proposed antenna
1

2

3

4

5

Number of parasitic elements, m

Fig. 6. Beamwidth in conical plane versus number of
director elements.

180(m = 3)20
F/B
15

Actual gain

10
5
0

0.01

0.1

1

10

ZL / Z0(a)

Directivity (dBi)

1.5
11.5
1
11
0.5
10.5
0.01

0.1
1
ZL / Z0(b)

100

Loss caused by termination load (dB)Actual gain (dBi), F/B (dB)

Fig. 4. Radiation pattern of unit array in xz-plane.

Fig. 5. Radiation properties versus load resistance:(a) Actual gain and F/B and (b) directivity andloss caused by termination load.

ing the termination condition so as to find their effecton F/B improvement. Figure 4 shows the radiation
pattern of a linear array with three parasitic elementsin the xz-plane for three termination conditions ofport #2: short, open, and ZL = Z 0. The backlobe has ahigh level when ZL is short or open and is reduced
only when ZL = Z 0. This is because the reflection of
the traveling wave at the terminated port is suppressed by the termination load. To find the optimum
termination condition, we calculated the relationshipbetween the radiation pattern and impedance ZL. FigVol. 5 No. 3 Mar. 2007

ure 5 shows the actual gain, F/B, directivity, and termination loss versus the termination load impedance
ZL, where the number of parasitic elements is three.
F/B is maximum when ZL is around Z 0. The improvement in F/B compared with the cases of short andopen termination is more than 8 dB. We also foundthat the directivity and termination loss are maximum
when ZL is around Z 0. By considering the slight gain
difference over the range of the termination load condition shown in this graph, we can explain this as follows. The load impedance ZL, which is almost thesame as the impedance of the feed line Z 0, does notattenuate the mainlobe radiation, but does attenuate
the undesired radiation caused by the reverse traveling wave, which comes up at the end of the line array.
From the above calculations, we found that applying
a termination load to the quiescent feed port is effective in enhancing F/B without incurring any reduction
in gain.
To design the sector antennas, we clarified the
beamwidth of the unit linear array. Numerical results
for conical-plane beamwidth versus the number ofparasitic elements are shown in Fig. 6, for the case inwhich port #2 is terminated by a resistor corresponding to ZL = Z 0 = 50 Ω. We found that the beamwidthof the linear array was narrower than that of the conventional array. This means that the element terminated by ZL also works as a director element. We also
found that the linear array with three parasitic elements has a beamwidth of almost 60°, making it suitable for a six-sector antenna.
3. Geometry and mechanism of six-sector
microstrip Yagi-Uda antenna with common
parasitic elements
The configuration of the six-sector microstrip YagiUda antenna on a circular substrate is shown in Fig.
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7. The substrate has ground metallization on the rearside. Three unit linear arrays intersect at the center at
an angle of 60°. Therefore, the shape of the central
parasitic element, which is shared by all three linear
arrays, is hexagonal. All of the parasitic elements areelectrically smaller than the exciting elements.
In the figure, c is the width of the central hexagonalparasitic element and c/a = 0.96. The diameter of thesubstrate is 1.83λ0 when the distance between theedge of the substrate and the center of the exciting
elements is set to λ0/4. The other dimensions andmaterial parameters shown in Fig. 7 are the same as
those described in Section 2. The substrate area isonly 25% of that of a conventional antenna with six
individual single-beam microstrip Yagi-Uda arrays ina radial configuration. It is also much smaller than theantennas that use the techniques described in refs. [9]and [10].
The key idea of this antenna is the radiation pattern
forming provided by controlling the port termination
condition at all quiescent elements. For example,
consider what happens when element #1 is excited
and all other elements are terminated and thus quiescent. Element #1 is strongly coupled to its adjacentparasitic element, the central common parasitic element, and finally to element #4. Therefore, this linear
array also works as a microstrip Yagi-Uda antenna,
whose beam direction is from #1 to #4. However, thislinear array is strongly coupled to the two other intersecting linear arrays, which causes undesired radiation. This undesired radiation can be suppressed bychanging the termination of the quiescent elements,
such as #2, #3, #5, and #6. This is discussed below.

4. Numerical analysis of six-sector antenna
4.1 Effect of coupling between intersecting
linear arrays
In the six-sector antenna shown in Fig. 7, there isstrong coupling among the three linear arrays, which
impacts the radiation pattern, because all the arrays
share the central parasitic element. The calculated
conical radiation pattern of the antenna in Fig. 7 isshown in Fig. 8. In this calculation, we treated thecase where all of the terminated ports are connected
to loads with a value of ZL = Z 0. Here, an infinite
ground plane and substrate are assumed to simplify
the calculation. It can be seen that the beamwidth ismuch larger than that of the single linear array. The
actual gain and F/B of the six-sector antenna arelower than those of the single linear array by 1 and 4
dB, respectively. This means that the two other linear
arrays resonate and generate undesired radiation,
which degrades the radiation pattern. The scatteringparameters of the six ports are listed in Table 1. The
amounts of power transmitted from #1 to #4 and from
#1 to #2 or #6 are almost the same, which indicatesthat the power fed to port #1 is distributed not only tothe excited linear array, but also to the intersecting
linear arrays at high levels. Moreover, we found thatthe total power consumed by the termination loads is
at least 20% of the fed power. Therefore, the gain and
φ
0
0

30

330

–10

(a)

300

|E θ | (dB)

–20

z
θ

240

120

x

φ

270

–30

150

y
#5

d′

h
#6

#2

c
g

d

g

a

a′

Table 1. Calculated S-parameters in crossed array antenna.
Feed port
Exciting element #1
Ground metallization

Fig. 7. Geometry of planar six-sector antenna.

76

(b)

Fig. 8. Radiation patterns in conical plane (θ = 45°):(a) Pattern of six-sector antenna with 50 Ω
termination at all ports and (b) pattern of single
linear array in Fig. 1(b) (m = 3).

#4
#3

210
180

S21

S31

S41

S51

S61

Amplitude (dB)

–10.97–15.75–10.80–15.75–10.97

Phase (deg.)

–95.5

76.0

44.0

76.0

–95.5

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Regular Papers

Z0
(1)
,
j tan βlwhere Z 0 is the characteristic impedance of the line,
which is equal to the impedance of the signal source,
Zin =

θ

z

Based on the design described above, we fabricateda six-sector antenna with 1.83λ0 and measured its
characteristics. Figure 11 shows the measured |S11 |,where ports #2, #3, #5, and #6 were open, port #4 wasconnected to a load of ZL = Z 0 = 50 Ω, and l = 0. Here,fc is the center frequency (5 GHz). The measured values generally agree with the calculated ones. The |S11 |was less than –15 dB in the 6.5% bandwidth region.
Resonance occurred at about f/ fc =1.11 due to the res60° Beamwidth20

Beamwidth
160
140
120
100
80
60
40
20

F/B

16
12
8
4

Actual gain
0

x

φ

y

5. Measurement at 5 GHz

0.51
βl / π1.5Beamwidth (deg.)

4.2 Radiation pattern improvement using
reactive termination load
To improve the radiation pattern, we examinedother termination conditions because the power consumed at the feed ports can be utilized by changing
the termination conditions. Termination with a purereactive load fully reflects the incident power at thefeed ports without any power consumption, and thereflection phase can be controlled by changing thereactance of the load. Arbitrary reactance values caneasily be obtained by attaching line stubs of various
lengths. A model of the six-sector antenna with lines
of length l is shown in Fig. 9. Each line has a singlepole three throw (SP3T) switch at its end and theswitch connects the end of the line to either a signal
source, resistive termination, or open termination.
When one port is excited, the port at the opposite endof the array is connected to the termination load to
reduce the backlobe, and all other ports are set toopen. To obtain a symmetric conical radiation patternwith controllable termination conditions, all lines
should be the same length. When the end of the line
is open, the line works as an open stub, and its impedance Zin observed at the port is expressed byand β is the phase constant in the line.
The calculated actual gain and beamwidth are
shown as functions of line length l in Fig. 10. For simplification, the substrate and ground metallization
were assumed to be infinite planes. The actual gain
and beamwidth exhibit periodicity because the reactance of the open line at the port also changes periodically, as can be seen from Eq. (1). Note that F/B andthe gain are maximum around βl = nπ (n = 0, 1, 2, ...)and the beamwidth is almost 60°, which is suitable
for a six-sector antenna. We found that making theline length an integral multiple of the half wavelength
gave the optimum radiation pattern in this six-sector
antenna geometry.

Actual gain (dBi), F/B (dB)F/B are strongly affected by the presence of the intersecting linear arrays.

2

Fig. 10. Actual gain, F/B, and conical-plane beamwidthversus line length l.

#4
#3

#5
6.5%

#6

#2

0

•

SP3T switch

••

•

••

l

Open termination

Signal source
Resistive termination (ZL)

Fig. 9. Circuit configuration with finite lines and SP3T
switches.

Vol. 5 No. 3 Mar. 2007

|S11| (dB)

#1
–10
–20
–30

(b)

(a)

–40
0.85 0.9 0.95 1 1.05 1.1 1.15
Frequency f /fc

Fig. 11. |S11| versus frequency in six-sector antenna:(a) Measurement and (b) calculation.

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Regular Papers

lobe direction. | Eθ | and | Eφ | are defined as the copolarization and cross-polarization components,
respectively. The radiation pattern of the single linear
array was also calculated to investigate the effect of
the intersecting arrays on the radiation performance.
For all the calculations with these models, we
assumed an identical finite ground plane. | Eφ | components are not shown in Figs. 13(b) and (c) because
no | Eφ | components appeared in the calculation.
The good agreement between the measured and calculated results confirms the validity of the numericaldesign. Moreover, the maximum cross-polarizationlevel was 10 dB lower than the co-polarization level.
The microstrip antenna’s cross-polarization component in the conical plane is not low if the substrate isplaced parallel to the xy-plane. In only the xz-plane, avery low level of | Eφ | component was observed when
the current direction on the microstrip antenna wasparallel to x-axis. Since the microstrip Yagi-Uda
array narrows the beam toward the xz-plane in thiscase, our antenna achieved a radiation pattern with arelatively low cross-polarization level. Furthermore,comparing the radiation pattern of our antenna with

3
2
(a)
(b)

1

0
0.85 0.9 0.95 1 1.05 1.1 1.15
Frequency f /fc

Fig. 12. Loss caused by termination versus frequency insix-sector antenna: (a) Measurement and (b)calculation.
z
0

(a)

30

30

| Eθ |, | E φ | (dB)

–10

0

90

–30

240

| Eθ |

x

120

0

90

–30

240

| Eθ |

120

30

30

60

–20
90

–30

240

150

210

180

0

–10

60

–20

150

210

30

30

θ

(c)

0

–10

60
| Eφ |

–20

θ

(b)

| Eθ | (dB)0

θ| Eθ | (dB)

Loss caused by termination load (dB)onance of the rectangular parasitic elements. The loss
caused by termination is shown in Fig. 12. The measured values also agree with the calculated ones. The
termination loss was less than 1 dB and this corresponds to the result shown in Fig. 5.
Figures 13 and 14 show, respectively, the verticalradiation pattern in the xz-plane and the conical radiation pattern at θ = 45°, which is equal to the main-| Eθ |

120
150

210

180

180

Fig. 13. Radiation patterns in vertical plane (φ = 0): (a) Measurement of six-sector antenna,(b) calculation of six-sector antenna, and (c) calculation of single linear array.

30

|Eθ |
330

| Eθ |, | E φ | (dB)

–10

0

–20
–30

270
|E φ |

120

240
210
180

0
30

|Eθ |
330

0

–20
|E φ |

120

270

240
150

210
180

0

|Eθ |

30

330

–10

300

–30

φ

(c)

–10

300

150

φ

(b)

| Eθ |, | E φ | (dB)00| Eθ |, | E φ | (dB)

φ

(a)

300

–20
–30

|E φ |

120

270

240
150

210
180

Fig. 14. Radiation patterns in conical plane (θ = 45°): (a) Measurement of six-sector antenna,(b) calculation of six-sector antenna, and (c) calculation of single linear array.

78

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Regular Papers

that of a single linear array, we did not observed any
increase in the cross-polarization component causedby the intersecting linear arrays. The reason ourantenna had low cross-polarization even with theintersecting linear arrays is that its geometry is symmetrical, which helps to cancel the radiation of thecross-polarization components from the intersecting
arrays. We found that we could obtain a conical-plane
beamwidth of 73° and F/B of 17 dB and that this sixsector antenna has a radiation pattern suitable for a
sector antenna.
The frequency dependence of the measured andcalculated F/B and actual gain are shown in Fig. 15.
For both F/B and actual gain values, the measured
and calculated values agree well. F/B exceeded 17 dB
and the gain exceeded 10 dBi around the center frequency. The actual gain of the six-sector antenna was
1 dB higher than that of the single linear array shownin Fig. 5. This is because of the difference in the
shape of the ground plane. In our analysis, we could
only treat infinitely planar substrates; such a substratecauses surface wave loss, which cannot occur in measurements. We think that the reason for the good
agreement between the calculated and measured gain
values is that the directivity enhancement provided by
the narrow beam and the radiation efficiency degradation happen to cancel each other out in the calculation. Meanwhile, dramatic F/B deterioration was seen
when f / fc > 1.01 and the frequency range for 1-dB
20

F/B (dB)

15
10
Measurement
Calculation

5

Actual gain (dBi)

0

14
12
10
8
6
4
2
0

0.96

0.98
1
1.02
Frequency (f /fc)
(a)

1.04

1 dB

gain attenuation was 0.955< f /fc < 1.017. That is, theactual available bandwidth of this antenna is about
5.5%, i.e., 0.955< f /fc < 1.01.
6. Antenna construction and measurement at 25
GHz
We investigated a WLAN system operating at 25
GHz [14] and found that using the sector antenna iseffective in maintaining the coverage area even when
the propagation loss is larger than that at 5 GHz. The
sector antenna design described in this paper provides
a sector antenna that can be easily configured to fit
within the width of a PC card. At a very high frequency, such as in the quasi-millimeter band, however, the manufacturing error will be serious because
the wavelength is short and the antenna properties arechanged by even small differences in antenna element
size. In this section, we present the measured performance of the antenna at 25 GHz and discuss the feasibility of this antenna.
A photograph of the fabricated six-sector antenna
for 25 GHz is shown in Fig. 16. The antenna diameter D is 22 mm and the substrate thickness h is 0.43
mm. As can be seen, the antenna’s small size makes
it suitable for a PC card terminal, for which the manufacturing error of the printed metal pattern on thesubstrate was less than 20 µm.
The measured scattering parameters of the fabricated antenna are shown in Fig. 17. Here, only port #1
was excited. The transmitted power was observed at
port #4, and the other ports were open. The calculated values, which were obtained using MoM, showthat |S11 | is less than –10 dB over the 13% frequency
range. The measured and calculated values of |S41 |mostly agree with each other. The power wasted at
the terminated port was 15% of the fed power.
Figure 18 shows the measured radiation patterns,
where the conical pattern was obtained at the angle ofθ = 45°, which corresponds to the peak direction of
Teflon substrate

Measurement
Calculation
0.96

0.98
1
1.02
Frequency (f /fc)
(b)

1.04

22 mm

Fig. 15. Frequency dependence of antennacharacteristics: (a) F/B and (b) actual gain.

Vol. 5 No. 3 Mar. 2007

Fig. 16. Photograph of the fabricated antenna for 25 GHz.

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0

Calculation

Relative power (dB)0
|S11| (dB)

–5
–10
–15
–20
–25

Measurement–30
0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2
Frequency (f /fc)(a)|S41| (dB)Calculation–10

Measurement
Calculation

–15
–20
–25
–30
–180 –120 –60 060
Direction θ °(a)0Measurement

Relative power (dB)

0
–5

–5

–10
–15
–20
–25
–30
0.8 0.85 0.9 0.95 1 1.05 1.1 1.15 1.2
Frequency (f /fc)(b)–5Calculation

120 180

Measurement

–10
–15
–20
–25
–30
–180 –120 –60 060
Direction φ °(b)

120 180

Fig. 17. Measured and calculated S-parameter:(a) |S11| and (b) |S41|.

Fig. 18. Measured and calculated radiation pattern:(a) Vertical plane (xz-plane ) and (b) conicalplane (θ = 45°).

the vertical plane radiation pattern. The measured andcalculated radiation patterns agree well in the verticalplane, and the mainlobe shape was predicted well by
calculation in the conical plane. There is a ripple inthe measured vertical radiation pattern, and we suspect that this is the effect of scattering by the feeding
cable. The measured peak gain of the antenna was
11.2 dBi, which is slightly higher than the result for 5
GHz. These results indicate that manufacturing error
is not a serious problem and support the feasibility of
our antenna.

at the quiescent ports that maximizes F/B and actual
gain. Based on our numerical design, we fabricated asix-sector antenna. Measured results show that it
offers a high F/B value of 17 dB, high actual gain of
at least 10 dBi, and a radiation pattern that is suitable
for a six-sector antenna. These results indicate that
our six-sector antenna offers both high radiation performance and a 75% reduction in substrate area compared with the equivalent conventional antenna with
six individual microstrip Yagi-Uda arrays in a radialconfiguration. We also fabricated a six-sector antenna for 25 GHz. Its diameter was 22 mm and the substrate thickness was 0.43 mm. This antenna achieved
a conical-plane beamwidth of 61° and actual gain of
11.2 dBi. These results indicate the feasibility of ourantenna for 25-GHz operation.

7. Conclusion
We are studying a novel compact planar six-sector
antenna that uses an array of dual-beam microstrip
Yagi-Uda antennas with common directors for use inmobile terminals. A numerical analysis of the unit
dual-beam array showed that terminating the quiescent ports with an appropriate load enhances the F/B
value by at least 8 dB. We investigated the radiation
pattern of a six-sector antenna configured by combining 3 dual-beam unit arrays. To reduce the effect ofundesired coupling between unit arrays, we connected the quiescent ports of the other two unit arrays tofinite-length lines that fully reflect the incident
power. We found the optimum termination condition
80

Acknowledgment
We thank Professor Dr. Kunio Sawaya and Associate Professor Dr. Qiang Chen of Tohoku University
and Professor Dr. Koichi Tsunekawa of Chubu University for their encouragement and advice.
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Naoki Honma
Research Engineer, Wireless Systems Innovation Laboratory, NTT Network Innovation Laboratories.
He received the B.E., M.E., and Ph.D degrees
in electrical engineering from Tohoku University, Miyagi, in 1996, 1998, and 2005, respectively. In 1998, he joined NTT Radio Communication Systems Laboratories. His current researchinterests include planar antennas for high-speed
wireless communication systems. He is a member of IEEE and the Institute of Electronics,
Information and Communication Engineers(IEICE) of Japan. He received the 2003 Young
Engineers Award from IEICE, the 2003 APMCBest Paper Award, and the 2006 IEICE Communications Society Best Paper Award.

Tomohiro Seki
Research Engineer, Wireless Systems Innovation Laboratory, NTT Network Innovation Laboratories.
He received the B.E., M.E., and Dr.Eng.
degrees in electrical engineering from Tokyo
University of Science, Tokyo, in 1991, 1993, and2006, respectively. Since joining NTT in 1993,
he has been engaged in research on planar antennas and active integrated antennas for the millimeter-wave and microwave bands. He is currently interested in system-in-package technologies for millimeter-wave communication systems. He is a member of IEEE and IEICE. He
received the 1999 Young Engineer Award fromIEICE and the 2006 IEICE Communications
Society Best Paper Award.

Kenjiro Nishikawa
Senior Research Engineer, Wireless Systems
Innovation Laboratory, NTT Network Innovation Laboratories.
He received the B.E. and M.E. degrees in welding engineering and the Dr.Eng. degree in communication engineering from Osaka University,
Osaka, in 1989, 1991, and 2004, respectively. In
1991, he joined NTT Radio Communication Systems Laboratories (now NTT Network Innovation Laboratories), where he has been engaged in
R&D of 3D and uniplanar MMICs on Si and
GaAs and their applications. He is currently
interested in millimeter-wave transceivers, system-in-package technologies, active integrated
antennas, and high-speed communication systems. He is a Technical Program Committee
Member of the IEEE Compound Semiconductor
IC Symposium, the IEEE Radio and WirelessSymposium, and the IEEE MTT-S International
Microwave Symposium. He served on the Steering Committee of the 2006 Asia-PacificMicrowave Conference. He is a member of IEEEand IEICE. He received the 1996 Young Engineer Award and the 2006 IEICE Communication
Society Best Paper Award from IEICE.

Shuji Kubota
Director, Vice President, Wireless Systems
Innovation Laboratory, NTT Network Innovation Laboratories.
He received the B.E. degree from the University of Electro-Communications, Tokyo, in 1980
and the Ph.D. degree in engineering from Osaka
University, Osaka, in 1995. Since joining Nippon
Telegraph and Telephone Public Corporation(now NTT) in 1980, he has been engaged in
R&D of forward error correction schemes andmodulation-demodulation schemes for satellite
and personal communication systems (PHS),software-defined radio systems, wireless LAN
systems, wireless access systems, and wide-area
sensor network systems. During 1991–1992, he
was a Visiting Researcher at the University ofCalifornia, Davis. He is a member of IEEE and
IEICE.

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