Efficiency and Damping Control Evaluation of aMatrix Converter with a Boost-up AC Chopper in
Adjustable Speed Drive System
Kazuhiro Koiwa and Jun-ichi Itoh
Department of Electrical Engineering
Nagaoka University of Technology
1603-1 Kamitomioka-machi Nagaoka, Japan
itoh@vos.nagaokaut.ac.jp
Abstract— This paper evaluates the damping control for amatrix converter (MC) configuration with a boost-up AC
chopper and the efficiency characteristics. This circuit
configuration can improve the voltage transfer ratio, but thelosses are increased due to the additional AC chopper. On theother hand, as a motor is driven at high rotating speed region,
the flux-weakening control is necessary for general MCs.
However, the copper loss is increased due to the fluxweakening control when an IPMSM is used. Therefore, thevalidity of the proposed circuit is evaluated in IPMSM
adjustable speed drive system in term of efficiency. In addition,
the proposed circuit is demonstrated by 3.7-kW prototype insimulation and experiments. As a result, it is revealed that theefficiency of the proposed system is higher when the motor
rated voltage is over 107% of the input voltage.

I.

INTRODUCTION

The matrix converters (MCs) which can convert an AC
power supply voltage directly into an AC output voltage ofvariable amplitude and frequency without the large energy
storages, such as electrolytic capacitors, have been studied
recently [1-3]. However, one of the disadvantages of the MC
is that the voltage transfer ratio, which defines as the ratio
between the output voltage and the input voltage, is being
constrained to 0.866. In terms of the motor drive system,
even a PM motor can be driven at rated frequency by an MCafter applying the flux-weakening, however, the output
current is increased and as a result the efficiency is reduced[3-4]. Therefore, in considering of widely applying the MCin the near future, the boost-up functionality is important.
In order to solve this problem, a Matrix-ReactanceFrequency Converter (MRFC), which consists of a MC andan AC chopper, has been studied [5]. Reference [5] showsthat the amplitude of the output voltage is controlled thevoltage which is higher than the input voltage. However the
MRFC requires many components including the boost-up
reactor and capacitor. In addition, the control of the MRFC
becomes complicated because the regular synchronizing
between the MC and the chopper is required.

The authors previously proposed a circuit topology whichconnects a V-connection AC chopper at the input stage ofthe MC that enables boost-up functionality in order toovercome the problem of voltage transfer ratio [6-8]. As aresult, it is not necessary to add the large inductor at primarystage of the AC chopper. Therefore, the V-connection AC
chopper and these components do not dominate the size andthe weight in comparison to the origin structure of a MC.
However, the proposed circuit has some problems asfollows: (i) the resonance occurs in the input filter, and (ii)
additional losses caused by the AC chopper. Thus, it isimportant to suppress the input filter resonance and toevaluate the total losses in comparison with that of theconventional MC.
In this paper, in order to suppress the input filter
resonance, the damping control is applied in either the AC
chopper or the MC. Three different types of the damping
controls are studied to apply in the proposed system, that is,(i) the current-type, (ii) the voltage-type and (iii) the outputside damping control. However, the characteristic of eachdamping controls has not been discussed and compared in
the adjustable speed drive system. On the other hand, thevalidity of the proposed system is evaluated in the adjustablespeed drive system in order to clarify the advantage of the
proposed system. Concretely, in order to drive the motor athigh speed, there are two possible solutions: (i) to apply theflux-weakening control or (ii) to boost-up the voltage by the
AC chopper. However, the method that can ensure highefficiency is not clear since the output voltage is limited.
Thus, in order to confirm the effectiveness of the proposedsystem, it is important to evaluate the total losses incomparison with the conventional MC when the boost-up
ratio is subjected to changes.
The constitutions of the paper are follows: first, the three
damping control methods are explained. Next, the losses in
the proposed circuit and the conventional MC are discussed
theoretically. Finally, the operation of the proposed system
will be demonstrated experimentally in a 3.7-kW Interior
Permanent Magnet Synchronous Motor (IPMSM). In

addition, the chopper and the copper losses are evaluated toclarify the validity of the proposed system.
II.

CIRCUIT TOPOROGY AND DAMPING CONTROLSTRATEGY

Fig. 1 shows the proposed circuit which connects a Vconnection AC chopper at the input stage of the MC. The
relationship between the input voltage vin and the output
voltage vout is expressed by




where, mc is the modulation index of the MC,  is theboost-up ratio of the chopper and vin is the input voltage.
Basically, the function of the AC chopper is simply boost-up
the input voltage. A feedback control is not required for thefilter capacitor since both the input and output sides of the
AC chopper are AC voltage. Therefore, the capacitor value is
not dominated by the voltage control response and thecurrent response of the input side in compared with a Backto-Back (BTB) system which consists of a PWM rectifier
and an inverter. As a result, the V-connection AC chopperand its components are not dominated by the volume and theweight in comparison to the origin structure of a MC.
However, the resonance between the input filter of the MC
and the AC chopper occurs in the proposed circuit. Therefore,
the damping control is needed to suppress the resonance.
Fig. 2 shows the control diagram of the three type
damping control methods for the proposed circuit. Several
studies have discussed and illustrated the features of each of
the damping controls [8-10]. Due to the differential incharacteristic, a suitable damping control is therefore
selected according to the application. Fig. 2(a) shows theblock diagram of the current-type damping control [8]. The
input currents are detected and treated as signals for thedamping control. Additionally, the high pass filter (HPF) is
used to decouple the oscillation components that are
produced from the input current. In the current-type, even theresonance distortion can be suppressed, current sensors arenecessary to implement physically due to the requirement of
current feedback. In contrast, Fig. 2(b) shows the control
diagram of the voltage-type damping control, where thecurrent sensors are not required [9]. The operation of thevoltage-type is follows: first, the capacitor current ic is
estimated from the differentiation of the capacitor voltage vc.
However, due to the noise disturbance, an HPF is used as a
pseudo differentiation. The cost of the voltage-type damping
control is cheaper than the current-type because theadditional sensors are not needed. Finally, Fig. 2(c) shows
the control diagram of the output damping control which isapplied in the MC [10]. The harmonic distortion in the input
stage can be suppressed by suppressing the harmonic
distortion in the output stage. This is because the input
instantaneous power is equaled to the output instantaneous
power in the MC.
As a result, the features of the damping control can besummarized as follows:(i) The current-type can achieve low

Figure 1. Circuit configuration of matrix converter with boost-up
chopper. In order to compensate output voltage, V-connection AC chopperis connected at input side.

1

mc

1

mc

Indirect
virtualcontrol [12]v out     mc  v in 

Figure 2. Control diagram of the damping controls. Current-type canachieve low THD. Voltage-type can suppress the resonance without
additional sensors. Output-side can simplify the overall control system.

harmonic components in the input current THD, (ii) thevoltage-type can effectively suppress the resonance without
additional sensors, which is benefits in terms of cost, and (iii)the output damping control can simplify the overall control
system. The effectiveness of the damping controls in termsof experimental will be continue to discuss in the chapter IV.
III.

DERIVATION METHOD OF CHOPPER LOSS AND
COPPER LOSS

A. Chopper loss
In order to consider the influence of the chopper loss forthe proposed system, the chopper losses such as theconduction loss and the switching loss are theoreticallyderived in this section.

Capacitor voltage vrs
Occurring period
Switching loss of Sc1, Sc4First, the conduction loss of Sc1 is expressed by
PconS1 

1 

v i dt 




ce r

0



0

where, vce is the on-state voltage of Sc1 which isexpressed by (3) and ir is the input current.

Input current iin

vce  k con1ir  k con 2 

0





Note that kcon1 and kcon2 are coefficient as defined fromthe on-state voltage characteristics in the datasheet of theswitching device. In this paper, SK80GM063 fromSEMIKRON is used as the switching devices in the AC
chopper. The rated voltage and the rated current of this IGBT
are 600V and 81A, respectively. From these equations, theconduction loss of Sc1 is expressed byPcon _ S 1 

k con1



2

I in 

2 2 k con 2

I in 

0

Occurring period
Switching loss of Sc2, Sc3/6/6

Figure 3. Integral period of switching loss of AC chopper. Switching loss
occurs during different period among each switching devices.
TABLE I.
CALCULATION PARAMETERS TO CALCULATE CHOPPER
LOSS. PARAMETERS OF SWITCHING DEVICE ARE OBTAINED FROMDATASHEET.

where, Iin is the effective value of the input current. In
similar, the conduction loss of Sc2 is expressed by (5).

Pcon _ S 2 ( 1)kcon1

2

I in 

2 2 (  1)kcon2





Iin 



Note that the equations of the conduction loss of Sc4 andSc3 are equal to (4) and (5), respectively.
Fig. 3 shows the integral period to calculate the switchingloss. The switching loss of the AC chopper is proportional tothe multiplication of the capacitor voltage vcrs and the input
current ir. Moreover, the integral period is different among
each of the switching devices. In other words, the integral
period for the Sc1 and Sc4 is to  from 5/6, and the integral
period for the Sc2 and Sc3 is to  from /6. Thus, theswitching loss Pton_s1 and Pton_s2 are expressed by
Pton _ S 1 Pton _ S 2 15 eon f s 6

2Vin f s(   1)[( 6  6) kton1 I in  12( 3  2)kton 2 ]24Vs1

Figure 4. Chopper loss which are conduction loss and switching loss, for
input current by calculation and simulation results.

 

vrs



dt
 e f

V
on

6



vrs
dt
Vs

eon  k ton1iin  k ton 2 ss

2  Vin f s[(5 6  3 2 )k ton1 I in  12(1  3 )k ton 2 ]24Vs

where, Vs is the nominal voltage when the switching loss
characteristics in the datasheet is used and fs is the switchingfrequency of the AC chopper. In addition, eon is theinstantaneous switching loss which is characterized by (8).







Note that kton1 and kton2 are the coefficients defined fromthe switching loss characteristics in the datasheet of theswitching device. Furthermore, the equations of theswitching loss of Sc4 and Sc3 are equal to (6) and (7).

In order to validate these equations, the calculation
results are compared with the simulation results which are
obtained from PSIM simulator. Table 1 shows thecalculation parameters. In the simulation, only the AC
chopper is simulated and current source is used as the load,in addition, the stray inductance and the stray capacitance are
not taken into the consideration.
Fig. 4 shows the conduction and switching losses of the
AC chopper by the theoretical calculation and the simulation[11]. Note that the chopper loss of the calculation results isthe sum of the equations of (4), (5), (6) and (7). As a result,the calculation results are well agreed with the simulation
results. Thus, the validity of the derived equations of the
chopper loss is confirmed. In other words, the chopper loss
can be easily calculated from the input current.

B. Copper loss by flux-weakening control
As an IPM motor is driven at high rotating speed region,
it is necessary to apply the flux-weakening control in the
motor control. However, the copper loss of the IPM motor isincreased because the d-axis current is flowed by the fluxweakening control. In this section, the control strategy of the
proposed system such as the flux-weakening control is
explained. Additionally, the equation of the copper loss
which is increased by the flux-weakening control, is
introduced from the d-axis and q-axis current. In order todrive an IPM motor in high rotating speed area within the
output voltage limitation of the MC, the flux-weakening
control is necessary to apply in the MC control. This isbecause the back electromotive force becomes higher withincreasing the rotating speed. The output voltage which isnecessary to control the motor can be degraded by the fluxweakening control. Therefore, the d-axis current id for theflux-weakening control is introduced.
Fig. 5 shows a vector diagram in the flux-weakening
control, where eq is the back electromotive force, v and v’ are
the terminal voltage in the IPM motor without/with the fluxweakening control, respectively. Note that the winding
resistance of the IPM motor Rw is not considered. The fluxweakening control in the IPM motor can equally weaken themagnetic flux in the permanent magnet from the d axis
armature magnetic flux. As a result, the rotation speed area
can be extended by implementing the flux-weakening control.
From Fig. 5, Vom which is the maximum the output phasevoltage of the MC, is expressed by (9).
Vom  v d  v q  ( Lq iq ) 2  ( Ld id  eq ) 2 
2

2



Therefore, the d-axis current in the flux-weakening
control at the maximum voltage is calculated by (10).

Figure 5. Vector diagram on flux-weakening control. In order to suppress
output voltage, d-axis current is flowed negatively. Motor can be driven athigh rotating speed by flux-weakening control.
IPM MOTOR PARAMETERS. COPPER LOSS IS CALCULATEDFROM THESE PARAMETERS.

TABLE II.

Rated mechanicalpower Pm

3.7 kW

Winding resistance Rw

Back electro-motive
force eq

151 V

Rated voltage Vn

180 V

d-axisinductance Ld
q-axisinductance Lq

Rated current In

14 A

Inertia moment J

0.0212 kgm2

Synchronous speed

1800 rpm

Number of pole pairs
Pn

3

s

Rated torque TeR

0.693
6.2 mH
15.3 mH

19.6 Nmconverter loss are increased. In order to validate the AC
chopper, it is evinced that the copper loss is lower in the
proposed system. In particular, previous equation has shownthat the copper loss is subjected to the input line voltage.
First, the torque of an IPM motor is expressed by (11).
T  Pn [eqiq  ( Ld  Lq )id iq ]







where, Pn is the number of pole pairs. From (11), thetorque of the IPM motor depends on the d-axis current andthe q-axis current. From (10) and (11), the d-axis current andq-axis current depend on Vom. Additionally, Vom is decided by
the input line voltage and voltage transfer ratio. In otherwords, in order to calculate the copper loss of the motor as
Vin is changed, the q-axis current is introduced for Vom. Thus,
Vom is expressed by (12) from (10) and (11).
Vom  [

Ld
T Lq
1
(

eq iq ) ]2  (Lq iq ) 2( Ld  Lq ) Pn Ld
iq





Note that the q-axis current should be obtained in orderto derive the copper loss. However, Eq. (12) cannot obtain
Ld

 
 the q-axis current. For this reason, the q-axis current iscalculated by numerical analysis. It should be noted that the
q-axis current is limited by (13).
The motor current is increased after implementing due tothe flux-weakening control, therefore the motor loss and the eq  Vom  (Lq i q ) 2
2

id 

iq 

Vom

Lq





On the other hand, the d-axis current can be calculated by(10). Therefore, the copper loss Pc is expressed by (14).
22

Pc  3Rw I o  3  Rw 

2(id  iq )
2





where, Rw is the primary winding resistance of the IPM
motor. Moreover, Io is the R.M.S. value of the output current.

C. Comparison with chopper loss to copper loss
Fig. 6 shows the calculation results which comparing
between the additional loss due to the AC chopper and theflux-weakening control. Note that the copper loss wascalculated from the motor parameter which is shown in
Table 2. Furthermore, additional copper loss in Fig. 6
subtracts copper loss due to the flux-weakening control
PC_CMC from that of the proposed system PC_BMC. The copper
loss of the proposed system does not depend on the input
voltage because the output voltage is constantly control at
180V by the AC chopper. On the other hand, when the input
voltage is degraded, the output voltage of the conventional
MC is decreased due to limited voltage transfer ratio. In
addition, the flux-weakening control in the IPM motor is notcapable of wide range application. Therefore, in order toextend the applications, the input voltage of the proposedsystem and the conventional MC are degraded. Then, highrotating speed of the IPM motor is possible to be simulated.
From the result, there is a range that additional copperloss(PC_CMC - PC_BMC) is negative. This is because the copper
loss in the proposed system is larger than that ofconventional MC. Additionally, in case that the ratio
between the motor rated voltage Vn and input voltage Vin(Vn/Vin) is less than 1.01 p.u., the chopper loss is larger thanthe copper loss. On the other hand, as Vn/Vin gets higher than
1.01 p.u., the chopper loss is lower than the copper loss. Thisis because the motor current and the copper loss are
increased due to the flux-weakening control when Vn/Vin ishigher.
Therefore, as the motor parameters and the switching
device characteristics are known, the effective region of the
AC chopper can be calculated by (4), (5), (6), (7), (10) and(12). In other words, it is not necessary to analyze the total
loss by the experiment. In terms of efficiency, effectivenessof the proposed system can be evaluated from theconventional MC by using the motor parameter and thedevice parameter without experiment.
IV.

EXPERIMENTAL RESULTS

A. Acceleration results of IPMSM
Fig. 7 shows the experimental acceleration tests for the
IPM motor which is driven by the proposed system with
vector control. Note that 4-step commutation sequence,
which uses the filter capacitor voltage, is applied to the

Figure 6. Loss comparison between chopper loss and copper loss of motorby calculation when input line voltage is changed. Copper loss in the figure
subtracts copper loss which occurs by the conventional MC PC_CMC fromthe copper loss by the proposed system PC_BMC.
TABLE III.

EXPERIMENTAL CONDITIONS.

Figure 7. Experimental acceleration test by MC with boost-up chopper.
Acceleration time is 0.8s. Boost-up AC chopper operates during rotating
speed is over 1560 r/min.

experiment. In Fig. 7, the AC chopper starts operating while
the rotating speed command * is over 1560rpm, and thenthe damping control is applied to suppress the input filter
resonance. Additionally, it is confirmed that the input and qaxis current are not drastically changed as the AC chopper isstarted operating. Therefore, the proposed system cancontinuously improve the voltage transfer ratio of the MC.
Fig. 8 shows the output line voltage characteristic basedon the output power of the converter. In this paper, the fluxweakening control is applied within rated voltage of the
motor when the load torque is changed at rated speed. Notethat the input line voltage is set to 180V. According to theresult, it is confirmed that the output line voltage is limited
up to 180V by the flux-weakening control. Thus, the fluxweakening control is validated by the experimental results.
When the load torque is changed at rated speed of the motor,
the flux-weakening control is applied at 1.3-kW output
power of the converter by the proposed system. On the other
hand, the flux-weakening control is started to apply at 400-W
output power of the converter. In other words, as the output
power of the converter is over 400-W load, the d-axis current
become larger due to the flux-weakening control by theconventional MC. As a result, the copper loss is increased
due to the flux-weakening control.

overshoots of the id and iq are larger than that of the case of
Fig. 9(a). The cause of the overshoot is to apply the output
damping control which compensates the transient componentof the id and iq. Next, Fig. 9(c) shows the torque transient
operation with the current-type damping control. From theresults, the convergence time of  is approximately 41.4ms,
which is similarly to Fig. 9(a). Finally, Fig. 9(d) shows thetorque transient operation with the voltage-type damping
control. Similarly, the convergence time of  isapproximately 42.8ms.

B. Evaluation of damping controls
Fig. 9(a) shows the torque transient operation without thedamping control. Note that the motor cannot be rotated dueto the harmonic distortion that is occurred in the case ofwithout the damping control when the load torque is changed
from 0% to 100%. Therefore, the torque step is degraded to
60% so that the motor can be rotated even the damping
control is not applied. As a result, the convergence time of
the rotating speed  is approximately 41.4ms. On the other
hand, Fig. 9(b) shows the torque transient operation with theoutput-side damping control. As a result, the convergencetime of  is approximately 46.2ms. Additionally, theFigure 8. Output line voltage characteristic with flux-weakening control.
Note that the input line voltage is set to 180V. As a result, the output line
voltage is limited up to 180V by the flux-weakening control.

Figure 9. Transient waveforms by each damping controls. As simulation time is 0.1s, load torque changes from 0% to 60%. The convergence time of  arefollows: (a) 41.4 ms, (b) 46.2 ms, (c) 41.4 ms and (d) 42.8 ms. The current-type is better than the other damping controls because the convergence time of 
is not changed by applying the damping control.

12 Without damping control

Voltage-type
damping control

108
6Current-type
4 damping control
200

Output-side
damping control

10 20 30 40 50 60 70 80 90 100
Output power of the motor (%)

Figure 10. Input current THD characteristics by without and with thedamping control. The input current THD is lower than 7% by the outputside damping control.

Figure 11. Converter efficiency and input power factor characteristics ofthe proposed system. Converter efficiency at maximum point is 94.8% at
2.3-kW output power.

Fig. 10 shows THD (Total Harmonic Distortion) of the
input current among the applied damping controls. From theresult, the input current THD can achieve lower than 7% atalmost all power ranges with the implementation of theoutput-side damping control. Accordingly, the damping
effect of the output-side is higher than that of others damping
controls.

C. Loss analysis of conventional MC and proposed system
In order to validate the proposed system in terms of theefficiency, the total loss characteristic of the proposedsystem is compared to the conventional MC by thesimulations and the experiments.
Fig. 11 shows the converter efficiency characteristic andthe input power factor characteristic, proportional to the
output power of the MC. First, it is confirmed that the unity
power factor can be obtained in more than 1-kW output
power. On the other hand, the converter efficiency at
maximum point is 94.8% at 2.3-kW output power.
Fig. 12 shows the d-axis current and the q-axis current
characteristics. Note that the experimental result was
obtained by the proposed system. In order to validate theequation of the copper loss, the calculation results arecompared to the experimental results. According to Fig. 12(a), the calculation result is almost agreed with theexperimental result. On the other hand, in Fig. 12-(b), theerror between the calculation result and the experiment result
is approximately 10%. This error is because the winding
resistance of the IPM motor is not considered in (13).
However, the trend of the calculation result almost agrees
with the experimental result. Thus, the validity of theequations for the copper loss can be confirmed. Besides, as
the ratio of Vn/Vin gets larger, the d-axis current becoming
larger. It means that the compensation value of the fluxweakening control is increased with the ratio of Vn/Vin. On
the other hand, the q-axis current is decreased as the ratio of
Vn/Vin is increased. According to (11), in case that the torqueis constant, the q-axis current is decreased when the d-axis
current is increased.

Figure 12. d-axis current and q-axis current comparison between
calculation and experimental results by the proposed system. Error
between calculation and experimental result is within 10%.

Fig. 13 shows the total loss comparison between the
proposed system and the conventional MC. Note that the
input voltage Vin was changed. Additionally, the iron loss
and the mechanical loss Pother were obtained from theexperiment by (16).

Pother  125  (Vn / Vin )  308  Accordingly, the total efficiency of the conventional MC
which is applying the flux-weakening control becomes low
as the input line voltage is decreased. This is because theconduction loss of the converter and the copper loss of theIPM motor are becoming larger due to the flux-weakeningcontrol. On the other hand, the AC chopper in the proposedsystem can easily improve the input line voltage which has
been decreased. For this reason, high efficiency of the
proposed system can be achieved in the adjustable drive
system.
According to the experimental result, when the ratio of
the input line voltage and motor rated voltage is over 1.07
p.u. , the total loss of the proposed system is less than that of
the conventional MC. However, in comparison with Fig. 6,
the turning point is different. The error between thecalculation and experimental result is approximately 5.6%.
The reason of this error is because the iron loss andmechanical loss is not considered in Fig. 6. The iron loss isdepending on the output line voltage and frequency.
Therefore, the validity of the proposed system is revealed as
the motor rated voltage is over 107% of the input voltage.
Fig. 14 shows the relationship between turning point of
Vn/Vin and the output power of the converter. As the result,
the turning point of Vn/Vin which becomes effectiveness ofthe proposed system, is degraded when the output power ofthe converter is increased. This is because the total efficiencyof the conventional MC is drastically degraded incomparison with the proposed system when the output powerof the converter increased. Thus, if the output power of theconverter are increased, effectiveness of the proposed system
can be expanded. In particular, it is revealed that theefficiency of the proposed system is higher than that of theconventional MC at the rated output power of the converter
when the motor rated voltage is over 90% of the input
voltage.
V.

CONCLUSION

In this paper, the validity of the proposed circuit isevaluated in term of the efficiency in an IPMSM drive
system. The proposed circuit was demonstrated by a 3.7-kW
IPMSM. As a result, it can be accomplished that the validityof the proposed system is confirmed as the motor rated
voltage is under 107% of the input voltage.
ACKNOWLEDGMENT
A part of this study was supported by Industrial
Technology Grant Program in 2009 from New Energy andIndustrial Technology Development Organization (NEDO)
of Japan.
REFERENCES
[1]

[2]

[3]

[4]

P. W. Wheeler, J. Rodriguez, J. C. Clare, L. Empringham: ”Matrix
Converters: A Technology Review” IEEE Transactions on Industry
Electronics Vol. 49, No. 2, pp274-288, 2002.
Q. Lei, F.Z.Peng, B. Ge: "Pulse-Width-Amplitude-Modulated
Voltage-Fed Quasi-Z-Source Direct Matrix Converter with maximum
constant boost", , Vol. , No. , pp. 641-646 (2012)
D. Casadei, M. Mengoni, G. Serra, A. Tani, L. Zarri: "Assessment ofan Induction Motor Drive for High Speed Operation based on Matrix
Converter", EPE2007, Vol. , No. , pp. (2007)
Ki-Chan Kim : "A Novel Magnetic Flux Weakening Method of
Permanent Magnet Synchronous Motor for Electric Vehicles", IEEE
Trans., Vol. 48, No. 11, pp. 4042 - 4045 (2012)

Turning point
Proposed circuit
ExperimentCalculation

540
520
500

Conventional MC
ExperimentCalculation

480
460
0.85

0.9
0.95
1
1.05 1.1
1.15
Motor rated voltage Vn / Input line voltage Vin

1.2

Figure 13. Total loss comparison between conventional MC and proposedsystem. In case that Vn/Vin is over 107%, total loss of the proposed system
is less than the conventional MC.

1.5
1.4
1.3
1.2
1.1
1
0.9
0.8
1

1.5
2
2.5
3
3.5
Output power of the motor Pm (kW)4

Figure 14. Relationship between the turning point of Vn/Vin and the output
power of the motor Pm by calculation result. The turning point of Vn/Vin is
the point that the total loss of the proposed system is lower than that of theconventional MC.
[5]

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