Input Impedance Matched AC-DC Converter
in Wireless Power Transfer for EV Charger
Keisuke Kusaka*, Jun-ichi Itoh** Nagaoka University of Technology, 1603-1 Kamitomioka Nagaoka Niigata, Japan

Abstract—This paper provides experimental verificationsof the AC-DC converter for an electrical vehicle (EV)battery charger which an input impedance is matched to acharacteristic impedance of a transmission line. In a highfrequency wireless power transfer system such as amagnetic resonant coupling (MRC), the input impedance of
the AC-DC converter in the receiving side of the wirelesspower transfer should be matched to the characteristic
impedance of the transmission line in order to suppress thereflected power. This paper presents the fundamental
characteristics of the AC-DC converter. The experimental
result shows that the AC-DC converter enables a conversion
from 13.56 MHz AC to DC with input impedance of
29.6+j0.51 Ω. Thus, the reflection coefficient is suppressedby 17.8 points compared with that of the conventional
capacitor input-type diode bridge rectifier with 25 Ω of aload resistance. The suppressed reflection coefficient can
reduce the reflection loss of the wireless power transfer
system.
Index Terms—Wireless power transfer, Impedance
matching, High-frequency, AC-DC converter

I. INTRODUCTION
In recent years, wireless power transfer methods suchas a magnetic induction for wireless power transfer andmicro-wave wireless power transfer are increasingly
studied [1-4]. In particular, a wireless power transfer
method using a magnetic resonant coupling (MRC),
which is reported in 2007, has been attracted incommunity [5-8]. The MRC has some advantages
compared with the conventional wireless power transfer
methods such as an electromagnetic induction and a
microwave power transfer methods. First, the MRC canbe operated in high efficiency in a middle-range
transmitting distance such as 1 m at high efficiency over
90% with reason of characteristic constructions oftransmitting coils. The transmitting efficiency is
proportional to product of the quality factor Q and acoupling coefficient k between the primary and secondary
coils. The transmitting coils have high quality factor Q.
The high quality factor provides the high efficiency in a
middle range transmission even when a coupling
coefficient is small value. Note that the coupling
coefficient decreases inversely with a cube of distance.
Second, the declination in transmission efficiency causedby position gap is relatively small. From abovementioned advantages, the MRC is found suitable toapply in battery chargers for electric vehicles (EV). The
transmitting coil and the receiving coil are planted on the

ground of parking areas, and behind of the EV
respectively. The batteries are charged automatically,
when the EV is parked at parking areas. Thus, thewireless power transfer system with MRC can improvethe conveniences of users.
In the wireless power transfer system with MRC, thesize of the transmitting coils depends on the transmitting
frequency. Considering the application for a EV battery
charger requires a wireless power transfer system tooperate in high-frequency close to 13.56 MHz in theindustry science medical (ISM) band because
transmitting coils are required being small size and lightweight. In this paper, 13.56 MHz is used as a transmitting
frequency.
AC-DC converters in the receiving side of the wirelesspower transfer system have to convert from 13.56 MHz
AC to DC due to the high-frequency transmission. In
addition, a reflected power occurs at boundary points of
the impedance in the high frequency region when
impedances are difference between the input impedanceof an AC-DC converter and characteristic impedance ofthe transmission line. In order to suppress the reflected
power between the transmission line and the AC-DC
converter, the input impedance of the AC-DC converter
requires to be matched to the characteristic impedance.
The reflected power reduces the efficiency of wirelesspower transfer.
In the previous studies, capacitor input-type dioderectifiers (CI-DBRs) that are performed as an AC-DC
converter in the receiving side have been tested [9].
However, when the CI-DBR is used in receiving side of
the high-frequency wireless power transfer system, alarge reflected power occurs because the CI-DBR cannot
control the input current and the input power factor. In
this scheme, the input current and the input power factor
depends on the load condition. Thus the input impedance
fails to match with the characteristic impedance of atransmission line.
This paper proposes an AC-DC converter which the
input impedance can be matched to the characteristic
impedance of the transmission line without highfrequency switching. The AC-DC converter can convert
from 13.56 MHz AC to DC with input impedance
matching. The input impedance matching reduce thereflection loss due to the reduction of the reflection power.
The first section in this paper describes the requiredconditions for input impedance matching of the AC-DC
converter. Secondly, a construction of the AC-DC

converter is presented. Thirdly, details of design methodof the AC-DC converter are provided. Finally, thesimulation results and experimental results are shown inorder to evaluate the validity of the proposed circuit.
II. INPUT IMPEDANCE MATCHING OF AC-DC CONVERTER
Generally, all of the impedance of the circuitcomponents including a high-frequency power supply, atransmission line and a load are matched to the one
characteristic impedance in high-frequency circuits. In
other words, the output impedance of the high-frequency
power supply, characteristic impedances of thetransmission line and input impedance of the load havesame impedance. In particular, the characteristic
impedance of 50 Ω is used widely because a transmission
loss has a minimum value when a polyethylene is used as
an insulating material in between an inner conductor andan outer conductor. Thus the 50 Ω is used throughout the
paper.
Assuming that the wireless power transfer system is
employed to battery charger for EV, a high-frequency ACshould be converted to a DC in the receiving side of thewireless power transfer system. So, an AC-DC converter
is required in the receiving side. As a result, the input
impedance of the AC-DC converter including thebatteries of the EV should be matched to thecharacteristic impedance in order to suppress the reflected
power.
Generally, a characteristic impedance of atransmission line, which is reference value for input
impedance of the AC-DC converter, does not include theimaginary part. It means that the 50 Ω of thecharacteristic impedance indicates the 50+j0 Ω. Therefore,
input voltage and current of the AC-DC converter should
be fill the following conditions.
(1) Vin / Iin = 50 Ω(2) Input power factor is 1 (cos θ = 1)
Where, Vin is the fundamental input voltage of the ACDC converter, Iin is the fundamental input current and θ isthe phase angle between the input voltage and the input
current. In a low-frequency region such as a commercial
frequency, power factor correction (PFC) circuits with a
PWM control are used widely. The PFC circuit enable
control the input current. In other word, the input
impedance of the PFC circuit can be controlled to thecharacteristic impedance. However, the PFC circuit
which is controlled by using the PWM control needs ahigh-frequency switching compared with an input
frequency. Thus, it is difficult to operate the conventional
PFC circuit with the PWM control when the input
frequency is constrained high-frequency such as 13.56
MHz by the wireless power transfer system. It is resultingthat the AC-DC converter without high-frequency
switching is required in the receiving side of the highfrequency wireless power transfer system.

III. INPUT IMPEDANCE MATCHED AC-DC CONVERTER
A. Circuit Configuration
Fig. 1 indicates the proposed input impedance
matched AC-DC converter. In the proposed circuit, a bidirectional boost chopper is connected at a subsequent
stage of a resonant-type rectifier shown in [10-11]. The
resonant-type rectifier enables the PFC operation withouthigh-frequency switching using a resonance between an
inductor and capacitors. When the load of the resonant
type-rectifier is light, the resonant frequency of the circuitis presented by (1) approximately. It means that the input
current flow in a path of L1, C1 and C2 for most of thetime in the input period.

f1 ≈1
2π LC 2

(1)

On the other hand, the resonant frequency closes in to(2) with increasing load because the resonant time decline.

f2 ≈

1
2π LC

(2)

Reference [11] used the resonant-type rectifier in thecommercial frequency. However, a low-frequency
operation of the resonant-type rectifier results in bulk of
circuit components; the resonant inductor and resonantcapacitors. In addition, the resonant capacitors shouldhave a high rated voltage because of series resonancebetween the resonant inductor and resonant capacitors.
Accordingly, the resonant-type rectifier is constrained touse electrolytic capacitors, which have large equivalent
series resistance, as a resonant capacitor in [11]. It causes
a decline of conversion efficiency of the resonant-type
rectifier.
The high-frequency such as 13.56 MHz operation of
the resonant-type rectifier can improve above-mentioned
disadvantages. An increasing of the input frequency
reduces the capacitances of the resonant capacitor. Hence,laminated ceramic capacitors which have low equivalent
series resistances can be used for resonant capacitors. For
this reason, a high power density and an improvement of
conversion efficiency can be achieved.
On the other hand, the resonant-type rectifier has aproblem with an input current control. The input
impedance has fluctuation due to the load condition in theresonant-type rectifier when the resistance load isconnected at subsequent stage. In order to solve theprevious problem, the bi-directional boost chopper with
input voltage control is connected at subsequent stage ofthe resonant-type rectifier. The input impedance matchingis satisfied regardless of the load condition because theresonant-type rectifier is operated at one operating pointdue to the voltage control by the bi-directional boost
chopper. Besides, the boost chopper can be operated at
low switching frequency compared with input frequency
because the role of the bi-directional boost chopper isstabilizing the output voltage of the resonant-type

rectifier. The voltage fluctuation of the rectifier output
voltage is enough small due to the large capacitance ofthe smoothing capacitor. For this reason, the highfrequency switching is not necessary for the boost
chopper. In this paper, the bi-directional boost chopper isswitched at 100 kHz. Note that, a laminated ceramic
capacitor which has low parasitic inductance in the highfrequency [12] is connected in parallel to the electrolytic
capacitor in order to improve a characteristic ofsmoothing capacitor in the high-frequency region because
the rectified voltage includes the second harmonics of the
input frequency of 27.12 MHz.
In the actual wireless power transfer system, the ACDC converter is connected to the receiving coil as a
power source. However, a RF power supply which has 50
Ω of output impedance is used for simplicity ofexperiments instead of the wireless power transfer system.
Similarly a stabilized power supply is used as a simulated
battery. In addition, SiC schottky barrier diodes (SiCSBD) are used in the resonant-type rectifier. The materialvalue of the SiC is larger than one of the Si in term of the
band-gap Eg and the breakdown field strength EB. So, themaximum operation frequency fmax of the 4H-SiC reach to
9.0 which is scaled based on the value of Si where themaximum operation frequency is an index ofperformances which is provided by material value [13].
The maximum operation frequency is defined by fmax ≈μEBEg0.5. Thus the SiC devices are suitable for highfrequency operation [14].

operating point of the resonant-type rectifier due to thecontrol of the rectifier output voltage vch. In the proposed
circuit, a fast dynamic response of the bi-directional boost
chopper is not necessary. So, the AVR control isconstructed by a PI control with 400 rad/s of naturalangular frequency. Also the ACR is constructed by a PI
control with 4000 rad/s in an inner loop of the AVR. The
input current from the rectifier irec is deal as a disturbance.

1
1 + sTiv

1
sCch

1
sLch

1
1 + sTic

Fig. 2. Control block diagram for S1 and S2 in the AC-DC
converter.

C. Parameters Design of the AC-DC converter
Fig. 3 presents the simulation results with input
impedance variation due to the circuit parameters;
inductance L and capacitance C when resonancefrequency which obtained by (1) are 16.37 MHz, 17.26
MHz and 19.23 MHz. The input impedance of theresonant-type rectifier which has the bi-directional boost
chopper at subsequent stage depends on the inductance Lof the inductor L1, capacitance C of C1 and C2, andrectifier output voltage vch. In this consideration, 223 V ofthe high-frequency input voltage and 500 V of therectifier output voltage is chosen because of assuming 1
kVA wireless power transfer system. Fig. 3 (a) provides
that input impedance is proportional to the inductance L.
Meanwhile, the resonant-type rectifier provides a high

60
50
40
30
20
10
Fig. 1. Proposed input impedance matched AC-DC converter for highfrequency wireless power transfer system.

0

0.2

0.4

0.6

0.8

1

0.8

1

(a) Absolute value of input impedance.

B. Control Method of Bi-directional Boost Chopper
Fig. 2 shows a control block diagram of the bidirectional boost chopper where Lch is the an inductanceof L2, Cch is the combined capacitance of C3 and C4, Tic is
the integral action time of an automatic current regulator(ACR) and Tiv is the integral action time of an automatic
voltage regulator (AVR). The operating point of theresonant-type rectifier is stabilized. This is attributed to
the ACR in chopper current ic and the AVR in therectifier output voltage vch of the bi-directional boost
chopper.
The bi-directional boost chopper stabilizes the

1
0.95
0.9
0.85
0.8

0

0.2

0.4

0.6

(b) Input power factor.
Fig. 3. Variation of input impedance by circuit parameters.

input power factor when the inductance is low. Contrary
to this, the input power factor is reduced progressively
with increasing inductance. Furthermore, the inductancehas to be designed enough large in order to ignore theeffect of parasitic inductance of the circuit for
implementation. The resonance frequency f1 of 19.23
MHz can satisfy the previous conditions. Thus, theresonant inductance L = 0.95 μH, the resonantcapacitance C = 0.14 nF are chosen in experiment andsimulation.
IV. SIMULATION RESULTS OF THE AC-DC CONVERTER
Fig. 4 presents the simulation results of the AC-DC
converter where the input voltage is 223 V, rated input
power is 1 kVA. The chopper current ich and the chopper
voltage vch track the referenced chopper current ich* andthe referenced rectifier output voltage vch* respectively.
The rectifier output voltage includes second harmonics
from the input frequency because the capacitor C3 ischarged by the rectifier current irec. The difference of the
impedance between the capacitor C3 and L2 allows therectifier current to flow into only the capacitor C3.
Focusing on the Fig. 4, the input current with asinusoidal waveform of the AC-DC converter is obtained
due to the rectifier output voltage control. Additionally,
the input power factor of the AC-DC converter close in to
1 of the fundamental input power factor approximately.
Incidentally, an absolute value of 51.9 Ω of the input
impedance |Żin| is calculated from simulation results; afundamental input voltage is 223V, a fundamental input
current is 4.29 A. The input impedance focuses attention
on an fundamental component because the wirelesspower transfer with high quality factor Q enable a powertransmitting without harmonics components.

Following previous statements, it is confirmed that the
input impedance of the AC-DC converter is 51.9+j0 Ω.
The input impedance includes an error of 1.9 Ω comparedwith the characteristic impedance of the coaxial cablesince the characteristic impedance of a coaxial cable is50+j0 Ω. However, an error of 1.9 Ω means that reflected
power occur against a travelling power at reflection
coefficient Γ = 1.8%. In other words, a 1.8 % oftravelling power which is outputted from high-frequency
power supply does not be reached to a load.
A reflected power with using a CI-DBR as an AC-DC
converter in the receiving side has fluctuation from
11.6% to 22.2% due to the load condition [9]. Hence, thereflected power can be suppressed up to 20.4 points withproposed AC-DC converter. The suppressing of reflected
power can reduce the reflection loss of the wireless powertransfer system.
V. EXPERIMENTAL RESULTS OF THE AC-DC CONVERTER
A. Experimental Setup
Experimental verifications are shown in this chapter.
Table 1 provides circuit parameters for the experimental
setup. The resonant capacitors are modified from 140 pF
to 150 pF for reason of convenience of procurement.
Moreover, the inductor L2 which is made by authors froman electrical steel sheet core. Note that the bi-directional
boost chopper in the proposed AC-DC converter isoperated in open-loop control for simplicity. The AC-DC
converter; especially the part of the resonant-type rectifier
is mounted on the printed circuit board (PCB) in order tocut down the effects of parasitic inductance and parasitic
resistance between the circuit components. In particular,
the parasitic inductance of a wire which is used in series
to resonant inductance affects the input impedance. For
this reason, the resonant-type rectifier should be mountedwith keeping in mind parasitic inductances.
Table 1. Circuit parameters and conditions for the experimental setup.
Items

Manufactures

Model number

Value

VishayIRFB11N50APBF

500 V
11 A

D1D4

Cree

C3D08060A

600 V
8A

L1

TDK

VLF10040T1R5N8R9(Remodeled)

950 nHMOSFET

S1
S2

Diode

Inductor

L2--

2.3 mHC1
C2

TDK

C3216C0G2J151JT

150 pFC3TDKCKG57NX7R2J474M

470 nF(in parallel)

C4nichicon

UPW2V221MRD

220 μF(in parallel)

C5

BHC
Components

ALS30A221DB450

220 μF

Capacitor

Fig. 4. Operation waveforms of the proposed AC-DC converter insimulation.

B. Experimental Waveforms
Fig. 5 (a) presents the operational waveforms of the
proposed AC-DC converter where the simulated battery

voltage VB is set to 90 V. Fig. 5 (b) provides theoperational waveforms with Low pass filters (LPF). The
input voltage and input current are indicated through LPF.
Note that the LPF which has cut-off frequency of 20
MHz is implemented in the oscilloscope. The
experimental results show that the input current achieves
a sinusoidal waveform with small distortion. Additionally,
the input current has similar characteristics to the result
which is obtained by using the simulation. Besides, DC
voltages of the rectifier output voltage vch and the output
voltage VB are obtained. It means that the AC-DC
converter enable a conversion from 13.56 MHz of AC to
DC without high-frequency switching. Note that, theinput voltage distortion results from noises which immix
into an input stage of the high-frequency power supply
which used as instead of output coils of wireless powertransfer system. From the experimental waveforms with
LPF shown in (b), the low-frequency components up to
20 MHz become sinusoidal waveforms roughly, and it
can be achieved unity input power factor.

shown in fig. 6 are scaled on the basis of amplitude at
13.56 MHz. Additionally, a sampling frequency of theoscilloscope is 1.25 GHz. Note that, the probes; adifferential probe (Tektronix, P5205) and a current probe(Tektronix, TCP305), which are used in theseexperiments, provide a limitation to the frequency
bandwidth at 100 MHz. For this reason a harmonics
component over 7th can be only become reference values.
The input current harmonics is suppressed less than -20
dB in the frequency bandwidth from 2nd to 20th. Thus,
an input current distortion (THD) is 11.2% (reference
value) in the bandwidth by 20th from a 13.56 MHz.
Focusing attention on the fundamental components of
the input voltage and current, the input voltage of 19.8 V
and input current of 0.67 A are obtained from harmonics
analysis. It shows that an absolute value of the input
impedance |Żin| is 29.6 Ω. Also, 4.1 deg of phasedifference between the input voltage and the input current
is obtained. The phase angle presents the input power
factor cos θ is 0.99. Hence, the input impedance of the
proposed AC-DC converter is 29.6+j0.51 Ω. The
experimental result has measurable error on the real partso the design value is 50+j0 Ω. An additional simulation
results confirmed that the error occur in reason ofparasitic capacitances of diodes on the upper arm in theresonant-type rectifier. A design considering the parasitic
capacitor can reduce the error.

0
-20
-40
(a)

Without filters.

-60
0
-20
-40
-60
1
(b) With filters (cut-off frequency is 20 MHz).
Fig. 5. Experimental waveforms of the proposed AC-DC converter.

C. Harmonics analysis
Fig. 6 indicates the harmonics analysis results of the
input voltage and input current of the AC-DC converterin experiments. The harmonic analysis is had to use withan oscilloscope (Tektronix, TDS5054B) in order toinspect experimental waveforms. The analysis results

3

5

7

9 11 13 15 17 19

Number of harmonics

Fig. 6. Harmonics analysis of input voltage and current.

D. Reflection CoefficientFig. 7 presents an experimental comparison between
the conventional CI-DBR with SiC-SBD and the
proposed AC-DC converter, in term of the reflection
coefficient. The reflection coefficient is defined from the
input impedances of each circuit. It means that thereflection coefficient is the proportion of reflection power
to travelling power. The reflection coefficient should bezero because the reflection power does not input into a
circuit. Note that the input impedance of the conventional
CI-DBR depends on the load condition. In other words
the fundamental harmonic input current depends on theload. Thus the reflection coefficients are measured ateach four different loads; 25 Ω, 33.3 Ω, 50 Ω and 100 Ω.
To put it in perspective, the reflection coefficient with theproposed AC-DC converter has minimum value
compared with using the conventional CI-DBR.
Especially, the reflection coefficient is suppressed by
17.8 points compared with the CI-DBR with the load
resistance 25 Ω. The experimental results show that the
AC-DC converter can suppress the reflection loss which
occurs at input side of the AC-DC converter in highfrequency wireless power transfer system. Furthermore, adesign considering the parasitic capacitances of diodes in
the resonant-type rectifier provides suppression of thereflection coefficient.

Fig. 7. Comparison of reflection coefficient between the CI-DBR andproposed AC-DC converter based on the experimental results.

VI. CONCLUSIONSIn this paper, the AC-DC converter which the input
impedance can match to the characteristic impedance wasproposed. The input impedance matching can reduce thereflection loss due to the reduction of the reflection power
which occurs due to the impedance miss-matching
between the input impedance of an AC-DC converter andcharacteristic impedance.
The experimental verifications confirmed that the ACDC converter enable a conversion from high-frequency
AC of 13.56 MHz to DC without high-frequency
switching with input impedance matching.
The input impedance of 29.6 + j0.51 Ω is obtained byexperiments. The experimental result has a measurable
error on the real part so the design value is 50+j0 Ω.
However, the reflection coefficient which indicates therate of generation of reflected power can be reduced by
17.8 points compared with using the conventional
capacitor input-type diode rectifier. Thus, it is confirmedthat the AC-DC converter is valid circuit configuration
for a receiving side of the high-frequency wireless powertransfer system. In addition the design considering theparasitic capacitance which is reason of error can reduce
the reflection coefficient more.
In the future work, the detail of the design method of
the proposed AC-DC converter will be optimized.

ACKNOWLEDGMENT
This research was partially supported by the Japan
Society for the Promotion of Science; Grand-in-Aid for
Scientific Research (B), 24360106, 2012-2015
REFERENCES[1] J. O. Mur-Miranda, G. Fanti, Y. Feng, K. Omanakuttan, R.
Ongie, A. Setjoadi, N. Sharpe: “Wireless Power Transfer
Using Weakly Coupled Magnetostatic Resonators, IEEE
EPE, pp. 4179-4186 (2010)[2] T. Fujita, Y. Kaneko, S. Abe: “Contactless Power Transfer
Systems using Series and Parallel Resonant Capacitors”,
IEEJ Trans. D, Vol. 127, No. 2, pp. 174-180 (2007) (inJapanese)[3] T. Horiuchi, K. Kawashima: “Study on Planar Antennas
for Wireless Power Transmission of Electric Vehicles”,
IEEJ Trans. D, Vol. 130, No. 12, pp. 1371-1377 (2010) (inJapanese)[4] E. Waffenschmidt, T. Staring: “Limitation of inductive
power transfer for consumer applications”, IEEE EPE, pp.
1-10 (2009)[5] A. Kurs, A. Karalis, R. Moffatt, J. D. Joannopoulos, P.
Fisher, M. Soljacic: “Wireless Power Transfer via Strongly
Coupled Magnetic Resonances”, Science, Vol. 317, No. ,pp. 83-86 (2007)[6] A. Karalis, J. D. Joannopoulos, M. Soljacic: “Efficient
Wireless non-radiative mid-range energy transfer”, Annalsof Physics, Vol. 323, No. 1, pp. 34-48 (2008)[7] S. Lee, R. D. Lorenz: “A Design Methodology for MultikW, Large Airgap, MHz Frequency, Wireless Power
Transfer Systems”, IEEE ECCE, pp. 3503-3510 (2011)[8] T. Imura, H. Okabe, T. Uchida, Y. Hori: “Study on Openand Short End Helical Antennas with Capacitor in Seriesof Wireless Power Transfer using Magnetic Resonant
Couplings”, IEEE The 35th IECON, pp. 3848-3853 (2009)[9] K. Kusaka, J. Itoh: “Experimental verification of rectifierswith SiC/GaN for wireless power transfer using a magnetic
resonance coupling”, IEEE The 9th PEDS, pp. 1094-1099(2011)[10] K. Matsui, I. Yamamoto, G. Erdong, M. Hasegawa, K.
Ando, F. Ueda, H. Mori: “A Novel Single-Phase Rectifier
Circuit to Generate High DC Voltage by LC Resonance in
Commercial Frequency”, IEEJ Trans. D, Vol. 127, No. 4,
pp. 368-374 (2007) (in Japanese)[11] I. Yamamoto, K. Matsui, U. Yao: “A Novel Single-Phase
Rectifier Circuit with Sinusoidal Input Current with LC
resonance in Commercial Frequency”, IEEJ Trans. D, Vol.
122, No. 8, pp. 838-844 (2002) (in Japanese)[12] H. Obara, M. Kamaga, T. Ito, Y. Sato: “An Investigationof Capacitors for Flying Capacitor Converters”, IEEJ
Trans. D, Vol. 131, No. 12, pp. 1393-1400 (2011) (inJapanese)[13] B. J. Baliga, M. Bhatnagar: “Comparison of 6H-SiC, 3CSiC, and Si for power devices”, IEEE Trans. On Electron
Devices, Vol. 40, No. 3, pp. 645-655 (1993)[14] M. Bakowski: “Status and Prospects of SiC Power
Devices”, IEEJ Trans. D, Vol. 126, No. 4, pp. 391-399(2006)