Improvement of Light Load Efficiency for BuckBoost DC-DC converter with ZVS using
Switched Auxiliary Inductors
Hayato Higa

Akira Sagawa

Jun-ichi Itoh

Dept. of Energy Environment Science
Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japanhhiga@stn.nagaokaut.ac.jp

Dept. of Electric Information
Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japansagawa_electro@stn.nagaokaut.ac.jp

Dept. of Electric Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japan
itoh@vos.nagaokaut.ac.jp

Abstract—This paper proposes a changing buck-boost
inductance for non-isolated bidirectional buck-boost DC-DC
converter with zero voltage switching (ZVS) modulation inorder to achieve high efficiency at wide load region and a widevoltage variation. In the proposed converter, the auxiliary
inductor and the bidirectional switch are connected in parallel
to a main inductor, and each connection is switched dependingon the load condition. At the light load region, the bi-directionalswitch is turned off for the reduction of the converter loss withthe large equivalent inductance. On the other hands, theauxiliary circuit is utilized at the heavy load region in order toextend the power translation range with the small equivalent
inductance. In addition, the sequence for the switching auxiliary
inductor is also proposed in order to prevent the surge voltageof the bidirectional switch and the DC offset current in theauxiliary inductor. From the experimental results, the root
mean square value of the inductor current is reduced by up to
23.8% compared with that of the conventional converter. In
addition, the validity of the proposed sequence is also confirmed,
e.g. no surge voltage at the turn off of the auxiliary inductor.
Moreover, 41% of the converter loss reduction at the light load
region.
Keywords— Bidirectional DC-DC converter; Zero voltage
switching; switching inductanceI.

INTRODUCTION

Recently, energy storage systems (ESS) have been appliedin DC micro-grid systems [1]-[3]. In ESS, a bidirectional
buck-boost DC-DC converter is generally required to obtain
the bidirectional operation regardless of DC-bus and battery
voltage conditions. In a power converter for ESS, the powerfluctuation due to renewable energy resources is compensated.
In addition, the power oscillation can vary over a wide rangeof level, whereas the battery voltage in ESS fluctuates due toa charge and discharge operation. Hence, it is cruciallyrequired for ESS to be able to achieve the high efficiency inwide load range and under any condition of battery voltage
variation [4]. However, the typical buck-boost converter
topology with a continuous current mode is the low overall
efficiency due to the switching loss of the hard switching
operation [5].
In order to reduce the switching loss, the soft switching
methods have been actively researched [6]-[8]. In thereference of [6], a zero voltage switching (ZVS) techniques
which utilize a resonance between additional components andthe junction capacitors have been proposed. In this method,
the duty ratio range is limited by the resonance period. Besides,a zero voltage transient (ZVT) circuit is also proposed in order978-1-5386-6054-6/18/$31.00 ©2018 IEEEto achieve ZVS with entire load region by [7], [8]. By thismethod, the inductor current flow is controlled in order todischarge the junction capacitor of the switching devices.
However, the additional conduction and the switching losses
occur in ZVT circuit, i.e. the lower converter efficiency.
As different approaches, there are several modulation
methods to achieve the soft switching [9-11], such as atriangular-current mode (TCM) applied for the achievementof ZVS [9], [10]. However, the converter efficiency at the lightload becomes lower due to the high current ripple. Meanwhile,the control method to achieve ZVS for a four-switch-buckboost DC-DC converter without additional components has
been proposed [11]. In this method, the current ripple isreduced compared to that of TCM. However, the current
ripple is still large in order to achieve ZVS over entire load.
Furthermore, there is tradeoff between the current ripple andthe high power capability. In other words, the problem in thismethod is low efficiency at the light load.
This paper proposes adding an auxiliary circuit with small
inductors in order to achieve the high efficiency in wide loadconditions including the high power capability. In particular,
the bidirectional switches and the auxiliary inductors areconnected parallel to the main inductor in order to reduce thecurrent ripple at the light load. The originality of this paper ischanging the equivalent inductance depending on the voltage
variation and the load condition. Note that the current ratingof the bi-directional switches and all inductors become small
because the auxiliary inductors are connected in parallel to themain inductor. Moreover, sequence method for the switching
auxiliary inductor is proposed to prevent the occurrence of
DC-offset current in the auxiliary-inductor current. This paper
is organized as follows; first, the circuit configuration and theoperation modes with the ZVS achievement are explained.
Second, the auxiliary circuit with the small inductors is
introduced. Third, the proposed switching sequence isintroduced. Finally, the effectiveness of the proposed circuitand sequence are confirmed by the experimental results.
II.

CIRCUIT CONFIGURATION AND OPERATION
PRINCIPLE

A. Circuit Configration
Figure 1 shows the circuit configuration of the four-switch
buck-boost converter with the switched auxiliary small
inductors. In order to minimize the inductors, the switchingfrequency is increased. This leads to the increase in theswitching loss due to the higher switching frequency. In orderto avoid this problem, the soft switching method is employed.
In addition, the auxiliary inductors are changed in accordancewith the output power.
B. Operation princeple
Figure 2 shows the bidirectional operation waveforms of
the switching period with the ZVS achievement [10]. In both
power flow operation, there are four modes in the switching
period. In order to achieve ZVS, the offset of the inductor
current I0 is maintained during the zero-current interval of aconventional discontinuous current mode. The offset current
for the ZVS achievement I0 is calculated by

I 0  Vin

Cds
Lall sin2Td
2Cds Lall



S1

SL1

S3

SL2

Lm

Vbat

L1

Vbus
L

S2

S4

iL

v1

v2

Fig. 1. Circuit configuration of bidirectional buck-boost
converter with switched auxiliary inductors. The DC bus voltage
varies from 300 V to 400 V, whereas the battery voltage changes
from 300 V to 350 V.

where Cds is the junction capacitor of MOSFET, Lall is thetotal inductance including the main inductance L and theauxiliary inductors.

1 Pv2 ( )iL ( ) d2 0
 2Vbat1 2  2SW Lall I 01  
Vbus



4SW Lall  Vbus 22  Vbat12  Vbat 22 





S1,S2
0

Charge operation

In addition, the transferred power is determined by thesecondary voltage v2 and the inductor current. Thus, thetransferred power is calculated by

S3,S4
0
iL
0
-I0

Next, the each switching timing1-3 is calculated by (3)-

S1,S2

(5).
1 



SW LI 0Vbus  Vbat2 3



2
Vbus
 VbusVbat  Vbat

2 



2 

Vbat
( 3  1 ) 
Vbus



Discharge operation

0
S3,S4
0
I0
0
iL

Mode 13 

SW Lall I 0 (Vbus  Vbat )VbusVbat   LI 2  4 L P SW
0
SW all ref
2
2
 
V

V
V

V

bus
bus
bat
bat
2
2
Vbus
Vbat







where sw is the switching angular frequency, Pref is thereference transferred power.
It should be noted that the total inductance Lall is decided
by the number of the auxiliary inductors. In the discharge
operation, the switching timing of S1 and S3 are switched. At
the condition of3=the maximum transferred power isachieved. The maximum average power Pmax can be calculated
by

2

1

0

Mode 2

3Mode 32
Mode 4

Fig. 2. Operation waveforms of switching period with ZVS
achievement. In this method, the offset current I0 is generated inorder to satisfy ZVS condition. In the discharge operation, theswitching timing of S1 and S3 are switched.
Pmax 



VbusVbat22
Vbus
 VbusVbat  Vbat



 V V  SW Lall 2
I 0  I 0 (Vbus  Vbat )  bus bat SW Lall 
 4

(6)

As shown in (6), the maximum transferred power which
can still achieve ZVS can be increased when the inductor
value becomes small.
Figure 3 shows the inductor current waveforms at the lightload condition. With the large inductance, the peak current canbe reduced at the light load compared to that of the small
inductor value. In addition, the offset current for ZVS I0 is also

Figure 4 shows the root mean square (RMS) current of theinductor current. RMS value of the inductor current isreduced by the large inductance. The problem with only one
main inductor is that the maximum power becomes small
when the inductance is large. In the conditions of the low
inductance, the current ripple at the light load increases, i.e.
the lower efficiency at the light load. In order to solve thistradeoff relationship between the power capability and thelight load efficiency, the inductance is changed in accordancewith the transferred power.
III.

Peak current
current isis reduced
Peak
max_L
IImax_L

iL :L
iL :L

Imax_2L

Imax_2L

iL :2L0iL :2L-I0
0

Fig. 1. Inductor current waveforms at light load when the-I0

inductance is changed. The peak current and RMS value are
reduced by increasing the inductance.
Fig. 3. Inductor current waveforms at light load when the
inductanceis changed. The peak current and RMS value are
5.0
reduced by increasing
theVinductance. L=268mH
Vin = 400
4.0
5.0

ILrms [A] ILrms [A]reduced by increasing inductance from (1). Figure 4 shows theroot mean square (RMS) current of the inductor current. RMS
value of the inductor current is reduced by the large
inductance. The problem with only one main inductor is thatthe maximum power becomes small when the inductance islarge. In the conditions of the low inductance, the current
ripple at the light load increases, i.e. the lower efficiency at thelight load. In order to solve this tradeoff relationship between
the power capability and the light load efficiency, theinductance is changed in accordance with the transferred
power.

SWITCHING SEQUENCE OF AUXILIARY
INDUCTOR

3.0
4.0

Vout = 300 V
Vin = 400 V
Vout = 300 V

2.0
3.0

L=268mH
L=357mH
Maximum value of transferred power

1.0
2.0

L=536mH
Maximum value of transferred power

0.0
1.0 0

At the switching auxiliary inductor, the DC offset current
might occur at the turn-on timing of the auxiliary switches.
Furthermore, the surge voltage occurs in accordance with theturn-off timing of the auxiliary inductor. In order to preventthe occurrence of the surge voltage and the DC-offset current,
the following switching sequence is proposed.

L=357mH

200
400
600
800
L=536mH
Transferred power [W]

1000

0.0
Fig. 2. RMS
inductor 400
current. At600
the light 800
load, RMS1000
value is0 value of200
reduced in proportion ofTransferred
the inductance.
The[W]
larger inductance leads
power
to the smaller transferred power. Therefore, the inductor value isFig. 4.
RMS value of inductor current. At the light load, RMS
changed by switching the auxiliary inductors.

value is reduced in proportion of the inductance. The larger
inductance leads to the smaller transferred power. Therefore, theinductor value is changed by switching the auxiliary inductors.

Figure 5 shows the switching sequence for the switching
auxiliary inductor. Fig. 5 (a) and (b) show the turn-on and the
Carrier of input side

Carrier of input side

Din
0
Carrier of output side

Carrier of output side

Dout
0

0

I0
0
0
-I0
H
SL1
L
H
SL2
L

SL1
SL2
Mode 1SL1SL2Mode 2

SL1

SL1

SL1

SL1

SL2

SL2

SL2

SL2

Mode 1 Mode 2 Mode 3

Mode 4

SL1 OFF SL1

ON

SL1

SL2 OFF SL2

ON

SL2
SL2
SL2
SL2
Mode 1 Mode 2 Mode 3 Mode 4

Mode 1Mode 2

SL1

SL1

SL1

(a) Turn on at discharge operation (b) Turn off at charge operation (c) Turn on at discharge operation (d) Turn off at discharge operation
Fig. 5. In the proposed switching sequence, the current detection is not required because the switching timing is synchronized to theswitching pattern of S1 to S4.

turn-off sequence at the charge operation, whereas Fig. 5 (c)and (d) show that at the discharge operation. In Fig. 5 (a) and(c), there is only one mode at the turn on of the bidirectionalswitch because the offset current is smaller than at theswitching timing of S1 and S3. In Fig. 5 (b) and (d), there are
four modes in the turn off of the auxiliary switches. In theMode 1 of the charge operation, SL1 and SL2 are on at the inputside carrier peak, whereas SL1 and SL2 are on at the output-side
carrier peak. In the mode 1 of Fig. 5 (b), SL1 and SL2 are on.
IV.

Table II Experimental conditions

EXPERIMENTAL RESULTS

A. Experimental conditions
A 1.0-kW prototype is tested in order to evaluate theproposed buck-boost converter with the switched auxiliary
inductor. Table I shows the experimental parameters. At theboth legs, IRFP460 (VISHAY) is selected. IRFP460(VISHAY) has the on-state resistance Ron = 270 mΩ, thevoltage rating Vrate = 500 V and the current rating Irate = 20 A.
Note that the minimum current for ZVS I0 is calculated with
about two times margin.

Element

SymbolValue

Rated power

Prated

1.0 kW

DC-bus voltage

Vbus

400 V, 300 V

Battery voltage

Vbat

300 V, 350 V

Dead time at HV side

Td

1 ms

Main inductor

L

532 mH

Auxiliary inductor

L1

532 mH

Swiching frequency

fsw

50 kHzMinimum current with L1

I0

1.5 A

Minimum current w/o L1

I0

1.7 A

IRFP460 (VISHAY)
Vrate=500 V, Irate=20 A, Ron=0.27 W

MOSFET

B. Steady state operation
Figure 6 shows the operation waveforms at the charge
operation. It should be noted that the experimental conditions
are the input voltage of 300 V and the output voltage of 350
V. Fig. 6 (a) shows that with auxiliary inductor, whereas Fig.
6 (b) shows that without auxiliary inductor. By applying themodulation for the ZVS achievement, the inductor current at
the switching timing is also larger than the minimum current
I0 at light load, i.e. ZVS achievement. In Fig. 6 (b), theminimum current for ZVS is reduced by the large inductance.
In Fig. 6, the RMS value with the auxiliary inductor is reduced
by 23.8% compared to that without the auxiliary inductor at
the same load.

v1
100 V/divv2
100 V/div0
iLRMS=2.1 A

4.3 A

iLRMS=1.6 A
3.0 A

Reduced by 23.8%

0Figure 7 shows the operation waveforms at the discharge
operation. Fig. 7 (a) shows that with the auxiliary inductor,
whereas Fig. 7 (b) shows that without the auxiliary inductor.
It should be noted that the experimental conditions are theinput voltage of 300 V and the output voltage of 350 V. In Fig.
7, the current direction is difference from Fig. 5, i.e. thechanging the power flow. By applying the modulation for the
ZVS achievement, the inductor current at the switching timingis also larger than the minimum current I0 at the light load. By
applying the switching auxiliary inductor, the inductor current
is reduced by 23.8%. In Fig. 7 (b), the minimum current for
ZVS I0 is also reduced by the large inductor value. The RMS
value with the auxiliary inductor is reduced by 18.2%.
C. Transient Response at Switching Axiliary Iductor
Figure 8 shows the transient waveforms of switching
auxiliary inductor at the discharge operation. In Fig. 8(a), theauxiliary inductor current is flown after the turn on of theauxiliary switch. In addition, it is confirmed that the DC offsetof the auxiliary-inductor current is only the offset current for
ZVS I0. In Fig. 8(b), the auxiliary inductor current has been
flown after the turn off of SL1. Then, SL2 is turned-off when
the auxiliary-inductor current becomes zero. Thus, the surge
voltage does not occur because the body diode of SL1 is turned
off naturally. Therefore, the low voltage rating device can beselected by applying the proposed switching sequence.
Figure 9 shows the transient waveforms of switching theauxiliary inductor at the discharge operation. Fig. 9 (a) showsv2
100 V/divv1
100 V/diviL 2 A/div
5 ms/diviL 2 A/div(a) With Laux(b) Without Laux

Fig. 6. Operation waveforms at charge operation. Without theauxiliary inductor, RMS value of the inductor current is reducedby 28.6% compared with the auxiliary inductor.
v2
100 V/divv1
100 V/divv2
100 V/div0
iLRMS=2.2 A

Reduced by 18.2%

v1
100 V/div

iLRMS=1.8 A

0iL 2 A/div

iL 2 A/div-4.6 A-3.3 A

5 ms/div(a) With Laux(b) Without Laux
Fig. 7. Operation waveforms at discharge charge operation. In
Fig. 7, the discharge operation is achieved because the directionof the inductor current is changed. Without the auxiliary
inductor, RMS value of the inductor current is reduced by 28.6%
compared with the auxiliary inductor.

Gate signal of SL1

iL 2 A/div
0

Gate signal of SL22 [ms/div]

0

0

5 ms

5 ms

0
Auxiliary inductor
current 2 A/div

Auxiliary inductor
current 1 A/div

Dorain source voltage
100 V/div

0

ZVS

0

(a) Turn on of Laux (b) Turn off of Laux
Fig. 8. Transient waveforms of switching auxiliary inductor
Laux at charge operation. In Fig. 8 (a), the DC offset current isonly the offset current for ZVS I0.
Gate signal of SL1

Gate source voltage
10 V/div(a) S1

2 [ms/div]iL 2 A/div
0

Gate signal of SL20

Dorain source voltage
100 V/div0Gate source voltage
10 V/div
ZVS

0
5 ms

Auxiliary inductor
current 2 A/div

Auxiliary inductor
current 1 A/div(a) Turn off of Laux(b) Turn off of Laux
Fig. 9. Transient waveforms of switching auxiliary inductor
Laux at discharge operation. In Fig. 9 (a), the DC offset current
is only the offset current for ZVS I0.
that at the turn on of the bidirectional switch, whereas Fig. 9(b)shows that at the turn off of the bidirectional switch. In Fig.
9(a) and (b), it is also confirmed that the DC offset of theauxiliary-inductor current is only the offset current for ZVS I0.
In addition, the auxiliary inductor current has been flown after
the turn off of SL1. Then, SL2 is turned-off when thebidirectional inductor current becomes zero. Thus, the surge
voltage does not occur because the body diode of SL1 is turned
off naturally.
D. ZVS operation
Figure 10 shows the operation waveforms of the gate
signal and the drain-source voltage. Fig. 10 (a) shows theoperation waveforms of S1, whereas Fig. 10 (b) shows that of
S4. In Fig. 8, ZVS is achieved because the main switch isturned on at zero voltage. In addition, the surge voltage does
not occur due to no recovery current at the both legs. In
addition, the turn off loss can be reduced by connecting asnubber capacitor in parallel to the main switches.
E. Efficiency characteristics
Figure 11 shows the efficiency characteristics
with/without the auxiliary inductor at the bidirectional
operation. Fig. 11 (a) shows the efficiency characteristics at
the input voltage of 400 V and the output voltage of 300 V,
whereas Fig. 11 (b) shows the efficiency characteristics at the
input voltage of 300 V and the output voltage of 350 V. In Fig.
11, the converter efficiency at the light-load range is improvedby the large inductance. In other words, the converter loss is(b) S4
Fig. 10. Operation waveforms of gate signal and drain-source
voltage. In Fig. 10 (a) and (b), ZVS is achieved by the offset
current for ZVS. In addition, the turn-off loss can be reduced byconnecting a snubber capacitor in parallel to the main switches.

reduced by up to 41%. In addition, the maximum efficiency of
98.7% is achieved as shown in Fig. 11 (a) of the light load. Bychanging the auxiliary inductor depending on the transferred
power, the high power capability is obtained. At the rated
power, the converter efficiency is 98.3% at the rated power.
Moreover, the high efficiency over the wide load range isachieved.
F. Load step response
Figure 12 shows the transient waveforms at the step-up
load and the step-down load. Fig. 12 (a) shows the step-up
load from 300 W to 500 W, whereas Fig. 12 (b) shows the
step-down load from 500 W to 300 W. In Fig. 12, the stablecurrent response is confirmed. In addition, the minimum
current for ZVS is still achieved the both step-up and the stepdown load, i.e. achievement of ZVS at the transient responseof the step-up and the step-down load.
V. CONCLUSION
This paper proposed the four-switch-buck-boost converterwith the switched auxiliary inductor in order to improve thelight load efficiency and achieve the high power capacity. In
the modulation method of the buck-boost converter, theinductor current including the offset current was applied inorder to achieve ZVS. In the proposed circuit, the auxiliary
inductors were switched in accordance with the transferred
power and the voltage conditions. By switching the auxiliary
inductor, the converter losses at light load was reduced. In
addition, the switching sequence for auxiliary inductor was
proposed. In the experiment, the validity of the proposed

99.0

Loss
-31%

With auxiliary
inductor

98.5

98.0

Efficiency [%]Efficiency [%]

98.5

99.0

With auxiliary
inductor

Without auxiliary
inductor

97.5
97.0

Loss
-41%

98.0

Without auxiliary
inductor

97.5
97.0

1p.u. 1 kW

1p.u. 1 kW
96.5

96.5

96.0
-1.0

-0.5

0.0

0.5

1.0

96.0
-1.0

-0.5

Input power [p.u.]

0.0
Input power [p.u.]

0.5

1.0

(a) Vin=400V, Vout=300V(b) Vin=300V, Vout=350V
Fig. 11. Efficiency characteristics with/without auxiliary inductor. By applying the large inductor value, the converter efficiency at thelight load is improved. In addition, the high power capability is obtained when the auxiliary inductor is active.
VLin 100V/div

VLout 100V/div

500 W

300 W

20 [ms/div]

VLin 100V/div

VLout 100V/div

300 W

500 W

20 [ms/div]00

iL 2A/diviL 5 A/div

iL 2A/diviL 5 A/div

0

Input power 100 W/div
500 W

300 W

0

Input power 100 W/div

5 [ms/div]

300 W

500 W

5 [ms/div](a) Reference power change: 500 W to 300 W(b) Reference power change: 300 W to 500 W
Fig. 12. Transient waveforms at step-up load and step-down load. In Fig. 12, the inductor current is seamlessly changed at the step-up andthe step-down load. In addition, the minimum current for ZVS is kept both the step-up and the step-down load.

method was confirmed by a 1.0-kW prototype. As results,
RMS value of the inductor current was reduced by up to
23.8%. In other words, the converter loss at the light load was
reduced by up to 41 % compared to that with the auxiliary
inductor. Therefore, the high efficiency over wide load wasachieved by switching the auxiliary inductor. In addition, theauxiliary inductor is smoothly changed by the proposed
switching sequence. In future work, the design method for theauxiliary inductor will be considered.
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