LT8365
Low IQ Boost/SEPIC/Inverting
Converter with 1.5A, 150V Switch
FEATURESDESCRIPTION

Wide Input Voltage Range: 2.8V to 60V
n Ultralow Quiescent Current and Low Ripple
Burst Mode® Operation: IQ = 9µA
n 1.5A, 150V Power Switch
n Positive or Negative Output Voltage Programmingwith a Single Feedback Pin
n Programmable Frequency (100kHz to 500kHz)n Synchronizable to an External Clockn Spread Spectrum Frequency Modulation for Low EMI
n BIAS Pin for Higher Efficiency
n Programmable Undervoltage Lockout (UVLO)
n Thermally Enhanced 16-lead MSOP Package
n AEC-Q100 Qualification in Progress

The LT®8365 is a current mode DC/DC converter with a
1.5A, 150V switch operating from a 2.8V to 60V input.
With a unique single feedback pin architecture it is capableof boost, SEPIC or inverting configurations. Burst Modeoperation consumes as low as 9µA quiescent current tomaintain high efficiency at very low output currents, while
keeping typical output ripple below 15mV.

n

An external compensation pin allows optimization of loop
bandwidth over a wide range of input and output voltages
and programmable switching frequencies between 100kHz
and 500kHz. A SYNC/MODE pin allows synchronization
to an external clock. It can also be used to select between
burst or pulse-skipping modes of operation with or without spread spectrum frequency modulation for low EMI.
For increased efficiency, a BIAS pin can accept a second
input to supply the INTVCC regulator. Additional features
include frequency foldback and programmable soft-start
to control inductor current during startup.

APPLICATIONS
Industrial and Automotive
Telecomn Medical Diagnostic Equipmentn Portable Electronicsnn

The LT8365 is available in a thermally enhanced 16-lead
MSOP package with four pins removed for high voltage
pin spacings.

All registered trademarks and trademarks are the property of their respective owners.

TYPICAL APPLICATION
400kHz, –250V Output Inverting Converter
0.1µF

VIN
9V TO 30V

20Ω

VOUT
–250V
10mA
0.22µF

0.1µF

10µH

Switching Waveforms

20Ω

10µF

IL
500mA/DIV

0.22µF
VIN

SW

EN/UVLO
LT8365
SYNC/MODE
RT

SS

1MΩ

FBX
BIAS

1µs/DIVINTVCC
GND

VC

8365 TA01b

3.24k
84.5k

107k

VOUT

VSW
50V/DIV

1µF

0.22µF
1nF

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100

8365 TA01a

Rev. 0Document Feedback

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1

LT8365
ABSOLUTE MAXIMUM RATINGSPIN CONFIGURATION(Note 1)

SW...........................................................................150V
VIN, EN/UVLO.............................................................60V
BIAS...........................................................................60V
EN/UVLO Pin Above VIN Pin, SYNC.............................6V
INTVCC ............................................................... (Note 2)
VC ................................................................................4V
FBX............................................................................±4V
Operating Junction Temperature (Note 3)
LT8365E............................................. –40°C to 125°C
LT8365J, LT8365H............................. –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C

TOP VIEWEN/UVLO 1
VIN 3
INTVCC 5NC 6
BIAS 7VC 8

16 SW1
17
PGND,
GND

14 SW21211109SYNC/MODE
SS
RT
FBX

MSE PACKAGE
VARIATION: MSE16 (12)16-LEAD PLASTIC MSOP
θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS PGND AND GND, MUST BE SOLDERED TO PCBORDER INFORMATIONLEAD FREE FINISH

TAPE AND REELPART MARKING*PACKAGE DESCRIPTIONTEMPERATURE RANGE

LT8365EMSE#PBF

LT8365EMSE#TRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 125°C

LT8365JMSE#PBF

LT8365JMSE#TRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 150°C

LT8365HMSE#PBF

LT8365HMSE#TRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 150°C

LT8365EMSE#WPBF

LT8365EMSE#WTRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 125°C

LT8365JMSE#WPBF

LT8365JMSE#WTRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 150°C

LT8365HMSE#WPBF

LT8365HMSE#WTRPBF

8365

16-Lead Plastic MSOP with 4 Pins Removed–40°C to 150°C

AUTOMOTIVE PRODUCTS**

Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix.
**Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. Thesemodels are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact yourlocal Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports forthese models.

Rev. 02

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LT8365
ELECTRICAL CHARACTERISTICS

The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, EN/UVLO = 12V unless otherwise noted.
PARAMETERCONDITIONSVIN Operating Voltage Range
VIN Quiescent Current at Shutdown

MIN

TYP

UNITS

60

V

l

1
1

2
15

μA
μA

l

2
2

5
25

μA
μA

l

9
9

15
30

μA
μA

l

1200
1200

1600
1850

µA
µA

l

22
22

40
65

µA
µA

4.4
4

4.65
4.25

V
V

l

2.8

MAX

VEN/UVLO = 0.2V
VEN/UVLO = 1.5V

VIN Quiescent Current
Sleep Mode (Not Switching)
Active Mode (Not Switching)

SYNC = 0V
SYNC = 0V or INTVCC, BIAS = 0V
SYNC = 0V or INTVCC, BIAS = 5V

BIAS Threshold

Rising, BIAS Can Supply INTVCC
Falling, BIAS Cannot Supply INTVCC

VIN Falling Threshold to Supply INTVCC

BIAS = 12V

BIAS – 2V

V

BIAS Falling Threshold to Supply INTVCC

VIN = 12V

VIN

V

FBX Regulation
1.568–0.822FBX Regulation Voltage

FBX > 0VFBX < 0VFBX Line Regulation

FBX > 0V, 2.8V < VIN < 60V
FBX < 0V, 2.8V < VIN < 60V

FBX Pin Current

FBX = 1.6V, –0.8V

l–10

RT = 432k
RT = 143k
RT = 84.5k

l
l
l

92
279
465

l
l

1.6
–0.80

1.636
–0.780

V
V

0.005
0.005

0.015
0.015

%/V
%/V

10

nA

113
321
535

kHz
kHz
kHz

Oscillator
Switching Frequency (fOSC)
SSFM Maximum Frequency Deviation(∆f/fOSC) • 100, RT = 143k

Minimum On-Time

Burst Mode, VIN = 24V (Note 6)
Pulse-Skipping Mode, VIN = 24V (Note 6)

Minimum Off-Time

14

l

100
300
500
20

28

%

110
110

200
200

ns
ns

100

115

ns

SYNC/Mode, Mode Thresholds (Note 5)

High (Rising), VIN = 24V
Low (Falling), VIN = 24V

l
l

0.14

1.3
0.2

1.7

V
V

SYNC/Mode, Clock Thresholds (Note 5)

Rising, VIN = 24V
Falling, VIN = 24V

l
l

0.4

1.3
0.8

1.7

V
V

fSYNC/fOSC Allowed Ratio

RT = 84.5k

0.95

1

1.25

kHz/kHzSYNC Pin Current

SYNC = 2V
SYNC = 0V, Current Out of Pin

10
10

25
25

µA
µA

2

2.7

A

SwitchMaximum Switch Current Limit Thresholdl

1.5

Switch Overcurrent Threshold

Discharges SS Pin

3

Switch RDS(ON)ISW = 0.5A

700

Switch Leakage Current

VSW = 150V

0.1

A
mΩ
1

µA

Rev. 0For more information www.analog.com

3LT8365
ELECTRICAL CHARACTERISTICS

The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, EN/UVLO = 12V unless otherwise noted.
PARAMETERCONDITIONS

MIN

TYP

MAX

UNITS

EN/UVLO Logic
EN/UVLO Pin Threshold (Rising)

Start Switching

l

1.576

1.68

1.90

V

EN/UVLO Pin Threshold (Falling)

Stop Switching

l

1.545

1.6

1.645

V

EN/UVLO Pin Current

VEN/UVLO = 1.6V

l

–50

50

nA

Soft-Start
Soft-Start Charge Current

SS = 0.5V

2µA

Soft-Start Pull-Down Resistance

Fault Condition, SS = 0.1V

220

Ω

Error Amplifier Transconductance

FBX = 1.6V
FBX = –0.8V

75
60

µA/V
µA/V

Error Amplifier Voltage Gain

FBX = 1.6V
FBX = –0.8V

185
145

V/V
V/V

Error Amplifier Max Source Current

VC = 1.1V, Current Out of Pin

7µA

Error Amplifier Max Sink Current

VC = 1.1V

7µA

Error Amplifier

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: INTVCC cannot be externally driven. No additional components or
loading is allowed on this pin.
Note 3: The LT8365E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT8365J and the LT8365H are guaranteed over the full –40°C to 150°C
operating junction temperature range.

Note 4: The IC includes overtemperature protection that is intended toprotect the device during overload conditions. Junction temperature will
exceed 150°C when overtemperature protection is active. Continuous
operation above the specified maximum operating junction temperature
will reduce lifetime.
Note 5: For SYNC/MODE inputs required to select modes of operation see
the Pin Functions and Applications Information sections.
Note 6: The IC is tested in a Boost converter configuration with the output
voltage programmed for 24V.

Rev. 04

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LT8365
TYPICAL PERFORMANCE CHARACTERISTICSFBX Positive Regulation Voltage
vs Temperature–0.7801.624–0.7851.616–0.790

1.608
1.600
1.592

–0.800–0.8051.576–0.8151.568–50 –25–0.820–50 –25

416

416

412

412

404
400
396
392
388
384

404
400
396
392
388

8365 G04

1.588365 G03

0

5 10 15 20 25 30 35 40 45 50 55 60
VIN (V)

2.4

140

2.3

130

1.9
1.8

120
110
100
90
80

1.7

70

1.6

60

80

100

VIN = 12V

100

75

50

25

0
–0.8

–0.4

0.0
0.4
0.8
VOLTAGE (V)50–50 –25

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)
8365 G08

8365 G07

1.2

1.6

Off-Time
Switch Minimum Off–Timevs Temperature

MINIMUM OFF TIME (ns)150

MINIMUM ON TIME (ns)

2.5

2.0

125

8365 G06

Switch Minimum On-Timevs Temperature

2.1

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)

8365 G05

Switch Current Limitvs Duty Cycle

2.2

EN/UVLO FALLING (TURN–OFF)

1.60

Normalized Switching Frequency
vs FBX Voltage

408

380

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)

4060
DUTY CYCLE (%)

1.62

1.54
–50 –25

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)

384

20

1.64

NORMALIZED SWITCHING FREQUENCY (%)

420

SWITCHING FREQUENCY (kHz)SWITCHING FREQUENCY (kHz)

420

0

1.66

Switching Frequency vs VIN

408

EN/UVLO RISING (TURN–ON)

1.68

8365 G02

Switching Frequency
vs Temperature

380–50 –25

1.70

1.56

8365 G01

SWITCH CURRENT LIMIT (A)

1.72–0.795–0.8101.5

1.74

VIN = 12V

1.584

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)

EN/UVLO Pin Thresholds
vs Temperature

EN/UVLO PIN VOLTAGE (V)

VIN = 12VFBX VOLTAGE (V)FBX VOLTAGE (V)1.632

FBX Negative Regulation Voltage
vs Temperature

130
120
110
100
90
80
70
60
50
40
30
20
10
0
–50 –25

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)
8365 G09

Rev. 0For more information www.analog.com

5LT8365
TYPICAL PERFORMANCE CHARACTERISTICS
VIN Pin Current (Active Mode,
Not Switching, Bias = 0V)
vs Temperature

VIN Pin Current (Sleep Mode, Not
Switching) vs Temperature30

2.0

VIN = 12V
1.8 VBIAS = 0V
V
= FLOAT
1.6 SYNC_MODE
VIN PIN CURRENT (mA)

VIN PIN CURRENT (µA)

VIN = 12V
27 V
BIAS = 0V
24 VSYNC_MODE = 0V
21
18
15
12
9

1.4
1.2
1.0
0.8
0.6

6

0.4

3

0.2

0
–50 –25

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)

0–50 –258365 G10

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)
8365 G11

VIN Pin Current (Active Mode, Not
Switching, Bias = 5V)
vs Temperature

Switching Waveforms(in DCM)50

VIN PIN CURRENT (µA)

VIN = 12V
46 VBIAS = 5V
VSYNC_MODE = FLOAT
42
IL
500mA/DIV

38
34
30
26

VSW
50V/DIV

22
18

1µs/DIV

14
10–50 –258365 G13

0 25 50 75 100 125 150 175
JUNCTION TEMPERATURE (°C)
8365 G12

Rev. 06

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LT8365
PIN FUNCTIONSEN/UVLO: Shutdown and Undervoltage Detect Pin. The
LT8365 is shut down when this pin is low and active
when this pin is high. Below an accurate 1.6V threshold,the part enters undervoltage lockout and stops switching.
This allows an undervoltage lockout (UVLO) threshold tobe programmed for system input voltage by resistively
dividing down system input voltage to the EN/UVLO pin.
An 80mV pin hysteresis ensures part switching resumes
when the pin exceeds 1.68V. EN/UVLO pin voltage below
0.2V reduces VIN current below 1µA. If shutdown and
UVLO features are not required, the pin can be tied directly
to system input.

RT: A resistor from this pin to the exposed pad GND copper (near FBX) programs switching frequency.

VIN: Input Supply. This pin must be locally bypassed. Be
sure to place the positive terminal of the input capacitor as
close as possible to the VIN pin, and the negative terminal
as close as possible to the exposed pad PGND copper(near EN/UVLO).
INTVCC: Regulated 3.8V Supply for Internal Loads. The
INTVCC pin must be bypassed with a 1µF low ESR ceramic
capacitor to GND. No additional components or loading isallowed on this pin. INTVCC draws power from the BIAS
pin if 4.4V ≤ BIAS ≤ VIN, otherwise INTVCC is powered by
the VIN pin.
NC: No Internal Connection. Leave this pin open.
BIAS: Second Input Supply for Powering INTVCC. Removes
the majority of INTVCC current from the VIN pin to improve
efficiency when 4.4V ≤ BIAS ≤ VIN. If unused, tie the pin
to GND.
VC: Error Amplifier Output Pin. Tie external compensation
network to this pin.
FBX: Voltage Regulation Feedback Pin for Positive or Negative Outputs. Connect this pin to a resistor divider between
the output and the exposed pad GND copper (near FBX).
FBX reduces the switching frequency during start-up andfault conditions when FBX is close to 0V.

SS: Soft-Start Pin. Connect a capacitor from this pin to
GND copper (near FBX) to control the ramp rate of inductor
current during converter start-up. SS pin charging current
is 2μA. An internal 220Ω MOSFET discharges this pin
during shutdown or fault conditions.
SYNC/MODE: This pin allows five selectable modes for
optimization of performance.
SYNC/MODE PIN INPUTCAPABLE MODE(S) OF OPERATION(1) GND or <0.14V

Burst(2) External Clock

Pulse-Skipping/Sync(3) 100k Resistor to GND

Burst/SSFM(4) Float (Pin Open)

Pulse-Skipping(5) INTVCC or >1.7V

Pulse-Skipping/SSFM

where the selectable modes of operation are:Burst = low IQ, low output ripple operation at light loads
Pulse-skipping= skipped pulse(s) at light load (aligned to clock)Sync = switching frequency synchronized to external clock
SSFM = Spread Spectrum Frequency Modulation for low EMI.
SW1, SW2 (SW): Output of the Internal Power Switch.
Minimize the metal trace area connected to these pins to
reduce EMI.
PGND,GND: Power Ground and Signal Ground for the
IC. The package has an exposed pad underneath the IC
which is the best path for heat out of the package. The pinshould be soldered to a continuous copper ground plane
under the device to reduce die temperature and increase
the power capability of the LT8365. Connect power groundcomponents to the exposed pad copper exiting near the EN/
UVLO and SW pins. Connect signal ground components tothe exposed pad copper exiting near the VC and FBX pins.

Rev. 0For more information www.analog.com

7LT8365
BLOCK DIAGRAM
L

VIN

VOUT
R3
OPT

R4
OPT

COUT

CIN
SW

EN/UVLOVIN

VBIAS (+)
VBIAS – 2V(–)

INTERNAL
REFERENCE
UVLO

+

SW1

+

+

–

–

SW2
BIAS

4.4V(+)
4.0V(–)

1.68V(+)
1.6V(–)

A6

UVLO

D

–

TJ > 170°C

3.8V REGULATORINTVCCINTVCCUVLO

SYNC/MODE

CVCC

RT
OSCILLATORFREQUENCYFOLDBACKERROR AMP
SELECT

FBX

1.6V

BURST
DETECT

ERROR
AMP

+

–

A1

–

+

R2

A7

A5
PWM
COMPARATOR

ERROR
AMP

OVERCURRENT

–

+

M1
DRIVER

A2

–

A3
ISS
2μA

UVLO
OVERCURRENT

M2

Q1

+
SLOPE

RSENSE

A4

–
SS

MAX
ILIMIT

+

–0.8V

1.5×
MAX
ILIMIT

+

VOUT

R1

SWITCH
LOGIC

SLOPE

–

R5

PGND/GND

VC
8365 BD

CSS

RC
CC

Rev. 08

For more information www.analog.com

LT8365
OPERATION
The LT8365 uses a fixed frequency, current mode control
scheme to provide excellent line and load regulation. Operation can be best understood by referring to the Block
Diagram. An oscillator (with frequency programmed by aresistor at the RT pin) turns on the internal power switch at
the beginning of each clock cycle. Current in the inductor
then increases until the current comparator trips and turns
off the power switch. The peak inductor current at which
the switch turns off is controlled by the voltage on the VC
pin. The error amplifier servos the VC pin by comparing thevoltage on the FBX pin with an internal reference voltage(1.60V or –0.80V, depending on the chosen topology).
When the load current increases it causes a reduction in
the FBX pin voltage relative to the internal reference. This
causes the error amplifier to increase the VC pin voltage
until the new load current is satisfied. In this manner, theerror amplifier sets the correct peak switch current levelto keep the output in regulation.
The LT8365 is capable of generating either a positive or
negative output voltage with a single FBX pin. It can be
configured as a boost or SEPIC converter to generate apositive output voltage, or as an inverting converter togenerate a negative output voltage. When configured asa Boost converter, as shown in the Block Diagram, the
FBX pin is pulled up to the internal bias voltage of 1.60V
by a voltage divider (R1 and R2) connected from VOUT
to GND. Amplifier A2 becomes inactive and amplifier A1
performs (inverting) amplification from FBX to VC. Whenthe LT8365 is in an inverting configuration, the FBX pin
is pulled down to –0.80V by a voltage divider from VOUT
to GND. Amplifier A1 becomes inactive and amplifier A2
performs (non-inverting) amplification from FBX to VC.

If the EN/UVLO pin voltage is below 1.6V, the LT8365
enters undervoltage lockout (UVLO), and stops switching.
When the EN/UVLO pin voltage is above 1.68V (typical),the LT8365 resumes switching. If the EN/UVLO pin voltage
is below 0.2V, the LT8365 draws less than 1µA from VIN.
For the SYNC/MODE pin tied to ground or <0.14V, theLT8365 will enter low output ripple Burst Mode operation for ultra low quiescent current during light loads tomaintain high efficiency. For a 100k resistor from SYNC/
MODE pin to GND, the LT8365 uses Burst Mode operation for improved efficiency at light loads but seamlesslytransitions to Spread-Spectrum Modulation of switchingfrequency for low EMI at heavy loads. For the SYNC/
MODE pin floating (left open), the LT8365 uses pulseskipping mode, at the expense of hundreds of microamps,
to maintain output voltage regulation at light loads byskipping switch pulses. For the SYNC/MODE pin tied to
INTVCC or >1.7V, the LT8365 uses pulse-skipping modeand performs Spread-Spectrum Modulation of switchingfrequency. For the SYNC/MODE pin driven by an external
clock, the converter switching frequency is synchronized
to that clock and pulse-skipping mode is also enabled. See
the Pin Functions section for SYNC/MODE pin.
The LT8365 includes a BIAS pin to improve efficiency
across all loads. The LT8365 intelligently chooses between
the VIN and BIAS pins to supply the INTVCC for best efficiency. The INTVCC supply current can be drawn fromthe BIAS pin instead of the VIN pin for 4.4V ≤ BIAS ≤ VIN.
Protection features ensure the immediate disable ofswitching and reset of the SS pin for any of the following
faults: internal reference UVLO, INTVCC UVLO, switch current > 1.5× maximum limit, EN/UVLO < 1.6V or junction
temperature > 170°C.

Rev. 0For more information www.analog.com

9LT8365APPLICATIONS INFORMATIONACHIEVING ULTRALOW QUIESCENT CURRENTTo enhance efficiency at light loads the LT8365 uses
a low ripple Burst Mode architecture. This keeps theoutput capacitor charged to the desired output voltage
while minimizing the input quiescent current and output
ripple. In Burst Mode operation, the LT8365 delivers single
small pulses of current to the output capacitor followed
by sleep periods where the output power is supplied by
the output capacitor. While in sleep mode, the LT8365
consumes only 9µA.
As the output load decreases, the frequency of single current pulses decreases (see Figure 1) and the percentage oftime the LT8365 is in sleep mode increases, resulting inmuch higher light load efficiency than for typical converters. To optimize the quiescent current performance at lightloads, the current in the feedback resistor divider must be
minimized as it appears to the output as load current. In addition, all possible leakage currents from the output should
also be minimized as they all add to the equivalent output
load. The largest contributor to leakage current can be dueto the reverse biased leakage of the Schottky diode (seeDiode Selection in the Applications Information section).
While in Burst Mode operation, the current limit of
the switch is approximately 325mA resulting in the
output voltage ripple shown in Figure 2. Increasing
the output capacitance will decrease the output ripple
proportionally. As the output load ramps upward fromzero the switching frequency will increase but only up
to the fixed frequency defined by the resistor at the

SWITCHING FREQUENCY (kHz)

600

VOUT
20mA/DIV

2µs/DIV

8365 F02

Figure 2. Burst Mode Operation

RT pin as shown in Figure 1. The output load at which
the LT8365 reaches the fixed frequency varies basedon input voltage, output voltage, and inductor choice.
PROGRAMMING INPUT TURN-ON AND TURN-OFF
THRESHOLDS WITH EN/UVLO PINThe EN/UVLO pin voltage controls whether the LT8365 isenabled or is in a shutdown state. A 1.6V reference and acomparator A6 with built-in hysteresis (typical 80mV) allow
the user to accurately program the system input voltage
at which the IC turns on and off (see the Block Diagram).
The typical input falling and rising threshold voltages canbe calculated by the following equations:
R3 + R4R4
R3 + R4
VIN(RISING, UVLO(+)) = 1.68 •R4
VIN(FALLING,UVLO(–)) = 1.60 •VIN current is reduced below 1µA when the EN/UVLO pin
voltage is less than 0.2V. The EN/UVLO pin can be connected directly to the input supply VIN for always-enabled
operation. A logic input can also control the EN/UVLO pin.
When operating in Burst Mode operation for light load
currents, the current through the R3 and R4 network caneasily be greater than the supply current consumed by theLT8365. Therefore, R3 and R4 should be large enough tominimize their effect on efficiency at light loads.

400

200

0

IL1+ IL2
100mA/DIV

VIN = 50V, VOUT = 100V, BOOST
0

10

20
30
40
LOAD CURRENT (mA)

50
8365 F01

Figure 1. Burst Frequency vs Load Current
Rev. 010

For more information www.analog.com

LT8365APPLICATIONS INFORMATIONINTVCC REGULATOR

Synchronization and Mode Selection

A low dropout (LDO) linear regulator, supplied from VIN,
produces a 3.2V supply at the INTVCC pin. A minimum
1µF low ESR ceramic capacitor must be used to bypass
the INTVCC pin to ground to supply the high transient currents required by the internal power MOSFET gate driver.

To select low ripple Burst Mode operation, for high efficiency at light loads, tie the SYNC/MODE pin below 0.14V(this can be ground or a logic low output).

No additional components or loading is allowed on thispin. The INTVCC rising threshold (to allow soft-start andswitching) is typically 2.65V. The INTVCC falling threshold(to stop switching and reset soft-start) is typically 2.5V.
To improve efficiency across all loads, the majority of
INTVCC current can be drawn from the BIAS pin (4.4V ≤BIAS ≤ VIN) instead of the VIN pin. For SEPIC applications
with VIN often greater than VOUT, the BIAS pin can be directly connected to VOUT. If the BIAS pin is connected toa supply other than VOUT, be sure to bypass the pin witha local ceramic capacitor.
Programming Switching Frequency
The LT8365 uses a constant frequency PWM architecturethat can be programmed to switch from 100kHz to 500kHz
by using a resistor tied from the RT pin to ground. A tableshowing the necessary RT value for a desired switchingfrequency is in Table 1.
The RT resistor required for a desired switching frequency
can be calculated using:
RT =

45.2k
fOSC1.009

where RT is in kΩ and fOSC is the desired switching frequency in kHz.
Table 1. SW Frequency vs RT ValuefOSC (kHz)RT (kΩ)

100

432

200

215

300

143

400

107

450

95.3

500

84.5

To synchronize the LT8365 oscillator to an external frequency connect a square wave (with 20% to 80% duty cycle)
to the SYNC pin. The square wave amplitude should havevalleys that are below 0.4V and peaks above 1.7V (up to
6V). The LT8365 will not enter Burst Mode operation at low
output loads while synchronized to an external clock, butinstead will pulse skip to maintain regulation. The LT8365
may be synchronized over a 100kHz to 500kHz range. The
RT resistor should be chosen to set the LT8365 switchingfrequency equal to or below the lowest synchronization
input. For example, if the synchronization signal will be
500kHz and higher, the RT should be selected for 500kHz.
For some applications it is desirable for the LT8365 tooperate in pulse-skipping mode, offering two major differences from Burst Mode operation. Firstly, the clock stays
awake at all times and all switching cycles are aligned tothe clock. Secondly, the full switching frequency is maintained at lower output load than in Burst Mode operation.
These two differences come at the expense of increased
quiescent current. To enable pulse-skipping mode, float
the SYNC pin.
To improve EMI/EMC, the LT8365 can provide spread
spectrum frequency modulation (SSFM). This feature varies
the clock with a triangle frequency modulation of 20%. For
example, if the LT8365's frequency was programmed toswitch at 500kHz, spread spectrum mode will modulate
the oscillator between 500kHz and 600kHz. The 20%
modulation will occur at a frequency: fOSC/256 where fOSC
is the switching frequency programmed using the RT pin.
The LT8365 can also be configured to operate in pulseskipping/SSFM mode by tying the SYNC/MODE pin above
1.7V. The LT8365 can also be configured for Burst Modeoperation at light loads (for improved efficiency) and
SSFM at heavy loads (for low EMI) by tying a 100k fromthe SYNC/MODE pin to GND.

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11LT8365APPLICATIONS INFORMATION
DUTY CYCLE CONSIDERATIONThe LT8365 minimum on-time, minimum off-time andswitching frequency (fOSC) define the allowable minimum
and maximum duty cycles of the converter (see Minimum
On-Time, Minimum Off-Time, and Switching Frequency
in the Electrical Characteristics table).
Minimum Allowable Duty Cycle =Minimum On-Time(MAX) • fOSC(MAX)Maximum Allowable Duty Cycle =1 – Minimum Off-Time(MAX) • fOSC(MAX)The required switch duty cycle range for a Boost converter
operating in continuous conduction mode (CCM) can becalculated as:
DMIN = 1 –DMAX = 1 –VIN(MAX)
VOUT + VD
VIN(MIN)
VOUT + VD

where VD is the diode forward voltage drop. If the above
duty cycle calculations for a given application violate
the minimum and/or maximum allowed duty cycles
for the LT8365, operation in discontinuous conduction
mode (DCM) might provide a solution. For the same VIN
and VOUT levels, operation in DCM does not demand as
low a duty cycle as in CCM. DCM also allows higher duty
cycle operation than CCM. The additional advantage of
DCM is the removal of the limitations to inductor value
and duty cycle required to avoid sub-harmonic oscillations
and the right half plane zero (RHPZ). While DCM provides
these benefits, the trade-off is higher inductor peak current, lower available output power and reduced efficiency.
SETTING THE OUTPUT VOLTAGE
The output voltage is programmed with a resistor divider
from the output to the FBX pin. Choose the resistor values
for a positive output voltage according to:⎛V⎞
R1 = R2 • ⎜ OUT – 1⎟
⎝ 1.60V ⎠Choose the resistor values for a negative output voltage
according to:⎛ |V | ⎞R1 = R2 • ⎜ OUT – 1⎟
⎝ 0.80V ⎠The locations of R1 and R2 are shown in the Block Diagram. 1% resistors are recommended to maintain output
voltage accuracy.
Higher-value FBX divider resistors result in the lowest
input quiescent current and highest light-load efficiency.
FBX divider resistors R1 and R2 are usually in the range
from 25k to 1M.
SOFT-START
The LT8365 contains several features to limit peak switch
currents and output voltage (VOUT) overshoot during
start-up or recovery from a fault condition. The primary
purpose of these features is to prevent damage to external
components or the load.
High peak switch currents during start-up may occur inswitching regulators. Since VOUT is far from its final value,
the feedback loop is saturated and the regulator tries tocharge the output capacitor as quickly as possible, resulting
in large peak currents. A large surge current may cause
inductor saturation or power switch failure.
The LT8365 addresses this mechanism with a programmable soft-start function. As shown in the Block Diagram,the soft-start function controls the ramp of the power switch
current by controlling the ramp of VC through Q1. This
allows the output capacitor to be charged gradually toward
its final value while limiting the start-up peak currents.
Figure 3 shows the output voltage and supply current for
the first page Typical Application. It can be seen that boththe output voltage and supply current come up gradually.
FAULT PROTECTION
An inductor overcurrent fault (> 3A) and/or INTVCC
undervoltage (INTVCC < 2.5V) and/or thermal lockout(TJ > 170°C) will immediately prevent switching, will
reset the SS pin and will pull down VC. Once all faults are
removed, the LT8365 will soft-start VC and hence inductor
peak current.
Rev. 012

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LT8365APPLICATIONS INFORMATION

VOUT
20V/DIVIL
200mA/DIV

10ms/DIV

8365 F03

Figure 3. Soft-Start Waveforms

FREQUENCY FOLDBACK
During start-up or fault conditions in which VOUT is very low,extremely small duty cycles may be required to maintaincontrol of inductor peak current. The minimum on-time
limitation of the power switch might prevent these low duty
cycles from being achievable. In this scenario inductor
current rise will exceed inductor current fall during eachcycle, causing inductor current to “walk up” beyond theswitch current limit. The LT8365 provides protection fromthis by folding back switching frequency whenever FBX
or SS pins are close to GND (low VOUT levels or start-up).
This frequency foldback provides a larger switch-off time,allowing inductor current to fall enough each cycle (seeNormalized Switching Frequency vs FBX Voltage in the
Typical Performance Characteristics section).
THERMAL LOCKOUT
If the LT8365 die temperature reaches 170°C (typical),the part will stop switching and go into thermal lockout.
When the die temperature has dropped by 5°C (nominal),
the part will resume switching with a soft-started inductor
peak current.

network is usually connected from the VC pin to GND.
The Block Diagram shows the typical VC compensation
network. For most applications, the capacitor should bein the range of 100pF to 10nF, and the resistor should
be in the range of 5k to 100k. A small capacitor is often
connected in parallel with the RC compensation network
to attenuate the VC voltage ripple induced from the output
voltage ripple through the internal error amplifier. The parallel capacitor usually ranges in value from 2.2pF to 22pF. A
practical approach to designing the compensation network
is to start with one of the circuits in this data sheet thatis similar to your application, and tune the compensation
network to optimize the performance. Stability should thenbe checked across all operating conditions, including load
current, input voltage and temperature. Application Note
76 is a good reference.
THERMAL CONSIDERATIONS

Care should be taken in the layout of the PCB to ensure goodheat sinking of the LT8365. Both packages have an exposed
pad underneath the IC which is the best path for heat outof the package. The exposed pad should be soldered to acontinuous copper ground plane under the device to reduce
die temperature and increase the power capability of theLT8365. The ground plane should be connected to large
copper layers to spread heat dissipated by the LT8365.
Power dissipation within the LT8365 (PDISS_LT8365) canbe estimated by subtracting the inductor and Schottky
diode power losses from the total power losses calculated
in an efficiency measurement. The junction temperatureof LT8365 can then be estimated by:		
TJ(LT8365) = TA + θ JA • PDISS_LT8365

APPLICATION CIRCUITSCOMPENSATION
Loop compensation determines the stability and transient
performance. The LT8365 uses current mode control toregulate the output which simplifies loop compensation.
The optimum values depend on the converter topology, thecomponent values and the operating conditions (including
the input voltage, load current, etc.). To compensate thefeedback loop of the LT8365, a series resistor-capacitor

The LT8365 can be configured for different topologies. The
first topology to be analyzed will be the boost converter,
followed by the SEPIC and inverting converters.
Boost Converter: Switch Duty Cycle
The LT8365 can be configured as a boost converter for theapplications where the converter output voltage is higher
than the input voltage. Remember that boost converters are
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13

LT8365APPLICATIONS INFORMATION
not short-circuit protected. Under a shorted output condition, the inductor current is limited only by the input supply
capability. For applications requiring a step-up converter
that is short-circuit protected, please refer to the Applications Information section covering SEPIC converters.
The conversion ratio as a function of duty cycle is:
VOUT
1
=
VIN
1− D

in continuous conduction mode (CCM).
For a boost converter operating in CCM, the duty cycleof the main switch can be calculated based on the output
voltage (VOUT) and the input voltage (VIN). The maximum
duty cycle (DMAX) occurs when the converter has theminimum input voltage:
DMAX =

VOUT − VIN(MIN)
VOUT

Discontinuous conduction mode (DCM) provides higher
conversion ratios at a given frequency at the cost of reduced efficiencies, higher switching currents, and lower
available output power.
Boost Converter: Maximum Output Current Capability
and Inductor Selection
For the boost topology, the maximum average inductor
current is:
I L(MAX)(AVG) = IO(MAX) •1
1•
1 − DMAXη

where η (< 1.0) is the converter efficiency.
Due to the current limit of its internal power switch, theLT8365 should be used in a boost converter whose maximum output current (IO(MAX)) is:I O(MAX) ≤VIN(MIN)
VOUT• (1.5A − 0.5 • ΔISW ) • η

Minimum possible inductor value and switching frequency
should also be considered since they will increase inductor
ripple current ∆ISW.
The inductor ripple current ∆ISW has a direct effect on the
choice of the inductor value and the converter’s maximum

14

output current capability. Choosing smaller values of∆ISW increases output current capability, but requires
large inductances and reduces the current loop gain (theconverter will approach voltage mode). Accepting larger
values of ∆ISW provides fast transient response andallows the use of low inductances, but results in higher
input current ripple and greater core losses, and reduces
output current capability. It is recommended to choose a∆ISW of approximately 0.60A.
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the inductor,
the inductor value of the boost converter can be determinedusing the following equation:
L =VIN(MIN)
ΔISW • fOSC

• DMAX

The peak inductor current is the switch current limit(maximum 2.7A), and the RMS inductor current is approximately equal to IL(MAX)(AVG).
Choose an inductor that can handle at least 2.7A without
saturating, and ensure that the inductor has a low DCR(copper-wire resistance) to minimize I2R power losses. Notethat in some applications, the current handling requirementsof the inductor can be lower, such as in the SEPIC topology
where each inductor only carries one-half of the total switch
current. For better efficiency, use similar valued inductors
with a larger volume. Many different sizes and shapes areavailable from various manufacturers (see Table 2). Choose acore material that has low losses at the programmed switching frequency, such as a ferrite core. The final value chosen
for the inductor should not allow peak inductor currents toexceed 1.5A in steady state at maximum load. Due to tolerances, be sure to account for minimum possible inductancevalue, switching frequency and converter efficiency.
For inductor current operation in CCM and duty cycles
above 50%, the LT8365's internal slope compensation prevents sub-harmonic oscillations provided theinductor value exceeds a minimum value given by:L>(2 • D – 1)
( –5 • D + 10 • D – 1) • (fOSC ) (1– D)
2

VIN

•

Lower L values are allowed if the inductor current operates
in DCM or duty cycle operation is below 50%.

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Rev. 0LT8365APPLICATIONS INFORMATIONtON

Table 2. Inductor Manufacturers
Sumida(847) 956-0666

www.sumida.com

TDK(847) 803-6100

www.tdk.com

Murata(714) 852-2001

www.murata.com

Coilcraft(847) 639-6400

www.coilcraft.com

Wurth(605) 886-4385

www.we-online.com

tOFF
ΔVCOUT

VOUT(AC)

RINGING DUE TO
TOTAL INDUCTANCE(BOARD + CAP)

ΔVESR

8365 F04

Figure 4. The Output Ripple Waveform of a Boost Converter

BOOST CONVERTER: INPUT CAPACITOR SELECTION
Bypass the input of the LT8365 circuit with a ceramic capacitor of X7R or X5R type placed as close as possible tothe VIN and GND pins. Y5V types have poor performanceover temperature and applied voltage, and should not be
used. A 4.7µF to 10µF ceramic capacitor is adequate to
bypass the LT8365 and will easily handle the ripple current. If the input power source has high impedance, orthere is significant inductance due to long wires or cables,
additional bulk capacitance may be necessary. This canbe provided with a low performance electrolytic capacitor.
A precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT8365.
A ceramic input capacitor combined with trace or cableinductance forms a high quality (under damped) tank circuit. If the LT8365 circuit is plugged into a live supply, theinput voltage can ring to twice its nominal value, possibly
exceeding the LT8365’s voltage rating. This situation iseasily avoided (see Application Note 88).
BOOST CONVERTER: OUTPUT CAPACITOR SELECTION
Low ESR (equivalent series resistance) capacitors should
be used at the output to minimize the output ripple voltage.
Multilayer ceramic capacitors are an excellent choice, as
they are small and have extremely low ESR. Use X5R or
X7R types. This choice will provide low output ripple andgood transient response. A 4.7µF to 47µF output capacitor
is sufficient for most applications, but systems with very
low output currents may need only a 1µF or 2.2µF output
capacitor. Solid tantalum or OS-CON capacitor can beused, but they will occupy more board area than a ceramic
and will have a higher ESR. Always use a capacitor with asufficient voltage rating. For High-Voltage DCM applications, smaller output capacitor values may be sufficient.

Contributions of ESR (equivalent series resistance), ESL(equivalent series inductance) and the bulk capacitancemust be considered when choosing the correct output
capacitors for a given output ripple voltage. The effect ofthese three parameters (ESR, ESL and bulk C) on the output
voltage ripple waveform for a typical boost converter isillustrated in Figure 4.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage ofthe output voltage), and how this ripple should be divided
between the ESR step ∆VESR and the charging/discharging ∆VCOUT. For the purpose of simplicity, we will choose
2% for the maximum output ripple, to be divided equally
between ∆VESR and ∆VCOUT. This percentage ripple willchange, depending on the requirements of the application,and the following equations can easily be modified. For a
1% contribution to the total ripple voltage, the ESR of theoutput capacitor can be determined using the following
equation:
ESRCOUT ≤0.01 • VOUT
ID(PEAK)For the bulk C component, which also contributes 1% to
the total ripple:
COUT ≥IO(MAX)
0.01 • VOUT • fOSC

The output capacitor in a boost regulator experiences highRMS ripple currents, as shown in Figure 4. The RMS ripple
current rating of the output capacitor can be determinedusing the following equation:
IRMS(COUT) ≥ IO(MAX) •DMAX
1 − DMAX
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15

LT8365APPLICATIONS INFORMATION
Multiple capacitors are often paralleled to meet ESR
requirements. Typically, once the ESR requirement issatisfied, the capacitance is adequate for filtering and has
the required RMS current rating. Additional ceramic capacitors in parallel are commonly used to reduce the effect ofparasitic inductance in the output capacitor, which reduceshigh frequency switching noise on the converter output.
CERAMIC CAPACITORS
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT8365 due to their piezoelectric nature.
When in Burst Mode operation, the LT8365’s switchingfrequency depends on the load current, and at very lightloads the LT8365 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT8365
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear.
If this is unacceptable, use a high performance tantalum
or electrolytic capacitor at the output. Low noise ceramic
capacitors are also available.

VIN

Table 3. Ceramic Capacitor Manufacturers
Taiyo Yuden(408) 573-4150

www.t-yuden.com

AVX(803) 448-9411

www.avxcorp.com

Murata(714) 852-2001

www.murata.com

BOOST CONVERTER: DIODE SELECTION
A Schottky diode is recommended for use with the LT8365.
Low leakage Schottky diodes are necessary when low
quiescent current is desired at low loads. The diode leakage
appears as an equivalent load at the output and should be
minimized. Choose Schottky diodes with sufficient reverse
voltage ratings for the target applications.
Table 4. Recommended Schottky Diodes

PART NUMBER

AVERAGEFORWARD REVERSE REVERSE
CURRENT VOLTAGE CURRENT
(A)
(V)
(µA)
MANUFACTURERDFLS1150

1

150

2

Diodes, Inc.

RB068L150TE25

2

150

3

ROHM

DFLS2100

2

100

1

Diodes, Inc.

VOUT

PGND

SW

°

1 EN

SW1 16PGND

3 VIN

SW2 14 SW

5 INTVCCSYNC 12

6 NC

SS 11

7 BIAS

GND

8 VC

RT 10
FBX 9
VOUT

°

8365 F05

Figure 5. Suggested Boost Converter Layout

Rev. 016

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LT8365APPLICATIONS INFORMATIONBOOST CONVERTER: LAYOUT HINTS
The high speed operation of the LT8365 demands carefulattention to board layout. Careless layout will result in performance degradation. Figure 5 shows the recommended
component placement for a boost converter. Note the vias
under the exposed pad. These should connect to a local
ground plane for better thermal performance.

The LT8365 can be configured as a SEPIC (single-ended
primary inductance converter), as shown in Figure 6. This
topology allows for the input to be higher, equal, or lower
than the desired output voltage. The conversion ratio asa function of duty cycle is:
VOUT + VD
D
=
VIN
1− D

D1

VIN

VOUT
CIN

L2
VIN

COUT

SW
LT8365

EN/UVLOINTVCCVOUT + VD
VIN(MIN) + VOUT + VD

Conversely, the minimum duty cycle (DMIN) occurs when
the converter operates at the maximum input voltage:
VOUT + VD
VIN(MAX) + VOUT + VD

Be sure to check that DMAX and DMIN obey:
DMAX < 1 – Minimum Off-Time(MAX) • fOSC(MAX)andDMIN > Minimum On-Time(MAX) • fOSC(MAX)where Minimum Off-Time, Minimum On-Time and fOSC
are specified in the Electrical Characteristics table.

in continuous conduction mode (CCM).
CDC

DMAX =DMIN =SEPIC CONVERTER APPLICATIONS

L1

The maximum duty cycle (DMAX) occurs when the converter
operates at the minimum input voltage:

SEPIC Converter: The Maximum Output Current
Capability and Inductor Selection
As shown in Figure 6, the SEPIC converter contains two
inductors: L1 and L2. L1 and L2 can be independent, but canalso be wound on the same core, since identical voltages
are applied to L1 and L2 throughout the switching cycle.
For the SEPIC topology, the current through L1 is theconverter input current. Based on the fact that, ideally, theoutput power is equal to the input power, the maximum
average inductor currents of L1 and L2 are:

FBX
GND

8365 F06

Figure 6. LT8365 Configured in a SEPIC Topology

In a SEPIC converter, no DC path exists between the input
and output. This is an advantage over the boost converter
for applications requiring the output to be disconnectedfrom the input source when the circuit is in shutdown.
SEPIC Converter: Switch Duty Cycle and Frequency
For a SEPIC converter operating in CCM, the duty cycleof the main switch can be calculated based on the output
voltage (VOUT), the input voltage (VIN) and the diodeforward voltage (VD).

IL1(MAX)(AVG) = IIN(MAX)(AVG) = IO(MAX) •DMAX
1 − DMAX

IL2(MAX)(AVG) = IO(MAX)In a SEPIC converter, the switch current is equal to IL1 +IL2 when the power switch is on, therefore, the maximum
average switch current is defined as:
ISW(MAX)(AVG) = IL1(MAX)(AVG) + IL2(MAX)(AVG)= IO(MAX) •1
1 − DMAX
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17LT8365APPLICATIONS INFORMATION
Given an operating input voltage range, and having chosen ripple current in the inductor, the inductor value (L1and L2 are independent) of the SEPIC converter can be
determined using the following equation:and the peak switch current is:
⎛
χ ⎞
1
ISW(PEAK) = ⎜1 + ⎟ • IO(MAX) •⎝
2 ⎠
1 − DMAXThe variable c in the preceding equations represents thepercentage peak-to-peak ripple current in the switch,
relative to ISW(MAX)(AVG), as shown in Figure 7. Then, theswitch ripple current ∆ISW can be calculated by:∆ISW = χ • ISW(MAX)(AVG)

The inductor ripple currents ∆IL1 and ∆IL2 are identical:∆IL1 = ∆IL2 = 0.5 • ∆ISW

L1 = L2 =ΔISW = χ • ISW(MAX)(AVG)

0.5 • ΔISW • fOSC

• DMAX

For most SEPIC applications, the equal inductor values
will fall in the range of 2.2µH to 100µH.
By making L1 = L2, and winding them on the same core, thevalue of inductance in the preceding equation is replacedby 2L, due to mutual inductance:
L=

ISW

VIN(MIN)

VIN(MIN)
ΔISW • fOSC

• DMAX

This maintains the same ripple current and energy storage
in the inductors. The peak inductor currents are:

ISW(MAX)(AVG)
t

DTS
TS

8365 F07

Figure 7. The Switch Current Waveform of the SEPIC Converter

The inductor ripple current has a direct effect on the
choice of the inductor value. Choosing smaller values of∆IL requires large inductances and reduces the current
loop gain (the converter will approach voltage mode).
Accepting larger values of ∆IL allows the use of low inductances, but results in higher input current ripple andgreater core losses. It is recommended that c falls in therange of 0.5 to 0.8.
Due to the current limit of its internal power switch, theLT8365 should be used in a SEPIC converter whose
maximum output current (IO(MAX)) is:
IO(MAX) < (1 – DMAX ) • (1.5A – 0.5 • ∆ISW ) • η

where η (< 1.0) is the converter efficiency. Minimum
possible inductor value and switching frequency should
also be considered since they will increase inductor ripple
current ∆ISW.

IL1(PEAK) = IL1(MAX) + 0.5 • ∆IL1
IL2(PEAK) = IL2(MAX) + 0.5 • ∆IL2

The maximum RMS inductor currents are approximately
equal to the maximum average inductor currents.
Based on the preceding equations, the user should choose
the inductors having sufficient saturation and RMS current ratings.
Similar to Boost converters, the SEPIC converter also needs
slope compensation to prevent subharmonic oscillations
while operating in CCM. The equation presented in the
Boost Converter section defines the minimum inductancevalue to avoid sub-harmonic oscillations when coupledinductors are used. For uncoupled inductors, the minimum
inductance requirement is doubled.
SEPIC Converter: Output Diode Selection
To maximize efficiency, a fast switching diode with a low
forward drop and low reverse leakage is desirable. The
average forward current in normal operation is equal tothe output current.

Rev. 018

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LT8365APPLICATIONS INFORMATION
It is recommended that the peak repetitive reverse voltage
rating VRRM is higher than VOUT + VIN(MAX) by a safetymargin (a 10V safety margin is usually sufficient).

VIN

+

+

L2

–

–

CIN

COUT

SW

The power dissipated by the diode is:

LT8365PD = IO(MAX) • VD

D1

GND

where VD is diode’s forward voltage drop, and the diodejunction temperature is:
TJ = TA + PD • RθJA

VOUT

+
+

8365 F10

Figure 8. A Simplified Inverting Converter

Inverting Converter: Switch Duty Cycle and Frequency

The RθJA used in this equation normally includes the RθJC
for the device, plus the thermal resistance from the board,
to the ambient temperature in the enclosure. TJ must notexceed the diode maximum junction temperature rating.
SEPIC Converter: Output and Input Capacitor Selection
The selections of the output and input capacitors of the
SEPIC converter are similar to those of the boost converter.
SEPIC Converter: Selecting the DC Coupling Capacitor
The DC voltage rating of the DC coupling capacitor (CDC,
as shown in Figure 6) should be larger than the maximum
input voltage:
VCDC > VIN(MAX)

CDC has nearly a rectangular current waveform. During theswitch off-time, the current through CDC is IIN, while approximately –IO flows during the on-time. The RMS rating of thecoupling capacitor is determined by the following equation:
IRMS(CDC) > IO(MAX) •

CDC

L1

For an inverting converter operating in CCM, the duty
cycle of the main switch can be calculated based on thenegative output voltage (VOUT) and the input voltage (VIN).
The maximum duty cycle (DMAX) occurs when the converter
has the minimum input voltage:
DMAX =

VOUT + VD
VOUT + VD + VIN(MIN)

Conversely, the minimum duty cycle (DMIN) occurs when
the converter operates at the maximum input voltage :DMIN =

VOUT + VD
VOUT + VD + VIN(MAX)Be sure to check that DMAX and DMIN obey :DMAX < 1 – Minimum Off-Time(MAX) • fOSC(MAX)andDMIN > Minimum On-Time(MAX) • fOSC(MAX)where Minimum Off-Time, Minimum On-Time and fOSC
are specified in the Electrical Characteristics table.

VOUT + VD
VIN(MIN)A low ESR and ESL, X5R or X7R ceramic capacitor works
well for CDC.
INVERTING CONVERTER APPLICATIONSThe LT8365 can be configured as a dual-inductor inverting
topology, as shown in Figure 8. The VOUT to VIN ratio is:

Inverting Converter: Inductor, Output Diode and Input
Capacitor Selections
The selections of the inductor, output diode and input
capacitor of an inverting converter are similar to those of
the SEPIC converter. Please refer to the corresponding
SEPIC Converter sections.

VOUT + VD
D
=
VIN
1− D
in continuous conduction mode (CCM).
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LT8365APPLICATIONS INFORMATION
Inverting Converter: Output Capacitor Selection

The RMS ripple current rating of the output capacitor
needs to be greater than:

The inverting converter requires much smaller output
capacitors than those of the boost, flyback and SEPIC
converters for similar output ripples. This is due to the fact
that, in the inverting converter, the inductor L2 is in series
with the output, and the ripple current flowing through theoutput capacitors are continuous. The output ripple voltage
is produced by the ripple current of L2 flowing throughthe ESR and bulk capacitance of the output capacitor:⎛⎞
1⎟
ΔVOUT(P–P) = ΔIL2 • ⎜⎜ESRCOUT +
⎟
8
•
f
•
C
OSC
OUT
⎝
⎠

IRMS(COUT) > 0.3 • ∆IL2

Inverting Converter: Selecting the DC Coupling
Capacitor
The DC voltage rating of the DC coupling capacitor (CDC,
as shown in Figure 8) should be larger than the maximum
input voltage minus the output voltage (negative voltage):
VCDC > VIN(MAX) + VOUT

CDC has nearly a rectangular current waveform. During
the switch off-time, the current through CDC is IIN, while
approximately –IO flows during the on-time. The RMS
rating of the coupling capacitor is determined by the following equation:

After specifying the maximum output ripple, the user canselect the output capacitors according to the preceding
equation.
The ESR can be minimized by using high quality X5R or X7R
dielectric ceramic capacitors. In many applications, ceramic
capacitors are sufficient to limit the output voltage ripple.

DMAX
1 − DMAX

IRMS(CDC) > IO(MAX) •A low ESR and ESL, X5R or X7R ceramic capacitor works
well for CDC.

TYPICAL APPLICATIONS
400kHz, 9V to 30V Input, –250V Output Inverting Converter
C9
0.1µF

D4

R6
20Ω

D3

C1
10µF

C7
0.22µFR5
D1 20Ω

SW
R1
1MΩ

EN/UVLO
LT8365
SYNC/MODE
RT

SS

R4
107k

FBX

VOUT

BIAS
INTVCC

GND
C2
0.22µF

VC
R3
84.5k

90
80

C5
0.22µF

D2
VIN

Efficiency
Efficiency
100

R2
3.24k

C4
1µF

EFFICIENCY (%)

VIN
9V TO 30V

C6
0.1µF

L1
10µH

VOUT
–250V
10mA

70
60
50
40
30
20
10
0

C3
1nF
8365 TA03a

0

1

2

3 4 5 6 7 8LOAD CURRENT (mA)

9 10 118365 TA03b

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
C6, C9: 12062C104KAT2A
C5, C7: CGJ5L3X7T2D224K160AA
Rev. 020

For more information www.analog.com

LT8365
TYPICAL APPLICATIONS
400kHz, 9V to 30V Input, –125V Output Inverting Converter

VIN
9V TO 30V

C6
0.1µF

L1
10µH
C1
10µF

D2

R5
20Ω

D1
VIN

R4
1MΩ

SW

VOUT
–125V
C5
0.22µF 15mA

EN/UVLOFBX

LT8365BIAS

SYNC/MODERT

INTVCC

SS

GND

VC
R2
84.5k

C2
0.22µF

R1
107k

R3
6.49k

C4
1µF

C3
1nF
8365 TA04a

D1, D2: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
C6: 12062C104KAT2A
C5, C7: CGJ5L3X7T2D224K160AA

400kHz, 9V to 30V Input, 250V Output Boost Converter
C7
0.1µF

D3
VOUT
250V
10mA

D2

VIN
9V TO 30V

L1
10µH

C6
0.22µF

R8
20Ω

D1

C5
0.22µF

C1
10µF

VIN

SW
R6
1MΩ

EN/UVLO
LT8365
SYNC/MODE
RT

SS

R3
107k

FBX

VOUT

BIAS
INTVCC

GND
C2
0.22µF

VC
R4
84.5k

R5
6.49k

C4
1µF

C3
1nF
8365TA05a

D1, D2, D3: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
C7: 12062C104KAT2A
C5, C6: CGJ5L3X7T2D224K160AA

Rev. 0For more information www.analog.com

21

LT8365
TYPICAL APPLICATIONS
400kHz, 9V to 30V Input, 125V Output Boost Converter
L1
10µH

VIN
9V TO 30V

D1

C1
10µF
VIN

C5
0.22µF

R4
1M

SW

VOUT
125V
20mA

EN/UVLOFBX

LT8365BIAS

SYNC/MODERT

INTVCC

SS

GND

VC
R2
84.5k

C2
0.22µF

R1
107k

R3
12.7k

C4
1µF

C3
1nF
8365 TA06a

D1: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
C5: CGJ5L3X7T2D224K160AA

400kHz, 4.5V to 60V Input, 48V Output SEPIC Converter
L1
47µH
C1
10µF

VIN

D1
L2
47µH

SW

C5
10µF

EN/UVLO
LT8365
SYNC/MODE
RT
R4
107k

GND
C2
10nF

Efficiency
Efficiency
100
90
80

FBX
BIAS

VOUT

INTVCC

SS

VOUT
48V
200mA at VIN = 12V
R1 250mA at VIN = 24V
1MΩ 300mA at V = 48V
IN

VC
R3
33k

R2
34.8k

C4
1µF

C3
6.8nF

EFFICIENCY (%)

VIN
4.5V TO 60V

C6
1µF

70
60
50
40
30
VIN = 12V
VIN = 24V
VIN = 48V

20
8365 TA07a

D1: DIODES INC. DFLS2100
L1: WURTH ELEKTRONIK 744873470
C5: MURATA GRM32ER71H106KA12L

10
0

0

0.05 0.10 0.15 0.20 0.25 0.30 0.35 0.40
LOAD CURRENT (A)8365 TA07b

Rev. 022

For more information www.analog.com

LT8365
TYPICAL APPLICATIONS
400kHz, 9V to 30V Input, 420V Output Boost Converter
C3
0.1µF

D5
VOUT
420V
4mA

D4
C2
0.1µF

C4
0.22µF

R2
20Ω

D3
D2

L1
10µH

VIN
9V TO 30V

C1
10µF

VIN

C5
0.22µF

R1
20Ω

D1

C6
0.22µF

SW
R3
1MΩ

EN/UVLOFBX

LT8365
SYNC/MODERT

INTVCC

SS

GND

VC

R4
3.83k

C7
1µF

R5
84.5k
C8
1nF

C9
0.22µF

R6
107k

VOUT

BIAS

8365 TA08a

D1, D2, D3, D4, D5: DIODES INC. DFLS1200
L1: WURTH ELEKTRONIK 74437324100
C2, C3: 12062C104KAT2A
C4, C5, C6: CGJ5L3X7T2D224K160AA

400kHz, 9V to 30V Input, –420V Output Boost Converter
C10
0.1µF

C9
0.1µF

VIN
9V TO 30V

C6
0.1µF

L1
10µH
C1
10µF

VIN

D1

D4

R5
20Ω

RT

SS

R4
107k

R7
20Ω

VOUT
–420V
4mA

C8
0.22µFC7
0.22µFD3

C5
0.22µF
R1
1MΩ

EN/UVLOSYNC/MODE

D5

D2

SW

LT8365

R6
20Ω

D6

FBX

VOUT

BIAS
INTVCC

GND
C2
0.22µF

VC
R3
84.5k
C3
1nF

C4
1µF

R2
1.91k

8365 TA09a

D1, D2, D3, D4, D5, D6: DIODES INC. DFLS1200
L1: WURTH ELEKTRONIK 74437324100
C6, C9, C10: 12062C104KAT2A
C5, C7, C8: CGJ5L3X7T2D224K160AA

Rev. 0For more information www.analog.com

23

LT8365
TYPICAL APPLICATIONS
400kHz, 10V to 30V Input, 210V/160V/110V Selectable, Low Output Ripple, HV Supply

VIN
10V TO 30V

C2
0.1µF

L1
10µH

C3
0.1µF

R10
47Ω

D1
D2

C1
10µF

C4
0.22µF

R1
20Ω

D3
D4

VIN

R3
1MΩ

EN/UVLOFBX

LT8365VOUT

BIAS

SYNC/MODE
RT

C9
0.1µF

C5
0.22µF

R2
20Ω

SW

VOUT
15mA

INTVCC

SS

GND

VC

C8
0.22µF

R9
107k

R7
27.4k

C6
1µF

R8
44.2k

R6
10.7k

R5
15.8k

R4
31.6k

210V

160V

110V

C7
1.8nF

NO JUMPER = 60V

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
L2: WURTH ELEKTRONIK 74479876147
C2, C3, C9: 12062C104KAT2A
C4, C5: CGJ5L3X7T2D224K160AA

8365 TA10a

400kHz, 10V to 30V Input, –210V/–110V Selectable, Low Output Ripple, HV Supply
C2
0.1µF

L1
10µH

VIN
10V TO 30V

C3
0.1µF

C1
10µF

D1

R1
20Ω

C4
0.22µF

D2
D3

R2
20Ω

SW
R3
1MΩ

EN/UVLOFBX

LT8365

SS

R7
107k

INTVCC
GND
C8
0.22µF

C9
0.1µF

VOUT

BIAS

SYNC/MODE
RT

VOUT
15mA

C5
0.22µF

D4
VIN

R8
47

VC
R6
44.2k

C6
1µF

R5
7.32k

C7
1.8nF

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
L2: WURTH ELEKTRONIK 74479876147
C2, C3, C9: 12062C104KAT2A
C4, C5: CGJ5L3X7T2D224K160AA

R4
8.06k
–210V
–110V

8365 TA11a

Rev. 024

For more information www.analog.com

LT8365
TYPICAL APPLICATIONS
Low IQ, Low EMI, 400kHz, –250V Output Inverting Converter with SSFM
C10
0.1µF

INPUT EMI FILTER
L2
470nH

VIN
12V

C1
4.7µF
×2
1210
50V

C11
0.1µF

L1
10µH
C2
10µF
50V
1206

C3
68µF
50V

VIN

RT

R1
100k

R5
1MΩ

GND
C4
0.22µF

R2
107k

VC

8365 TA12a

AVERAGE CONDUCTED EMI (dBµV)

PEAK CONDUCTED EMI (dBµV)

Conducted EMI Performance (CISPR25 Class 5 Average)
50

60
50
40
30
20
10
CLASS 5 PEAK LIMITMEASURED EMISSIONSAMBIENT NOISE
1FREQUENCY (MHz)1012V INPUT TO –250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

40
30
20
10
0
–10
–20

–40
0.1

30

30
8365 TA12d

Radiated EMI Performance

60

50

50

AVERAGE RADIATED EMI (dBµV/m)

PEAK RADIATED EMI (dBµV/m)

10

Radiated EMI Performance(CISPR25
Class 5 Average)(CISPR25 Class
5 Average)

60

40
30
20
10
CLASS 5 PEAK LIMITMEASURED EMISSIONSAMBIENT NOISE
100

1
FREQUENCY (MHz)12V INPUT TO –250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

8365 TA12b

Radiated EMI Performance

30

CLASS 5 AVERAGE LIMITMEASURED EMISSIONSAMBIENT NOISE–30

Radiated EMI (CISPR25
Performance(CISPR25Class
5 Peak) Class 5 Peak)

–20

R4
3.24k

C6
1µF

R3
84.5k
C5
1nF

70

–10

VOUT

INTVCC

SS

60

0

C8
1nF
500V
1206

BIAS

Conducted EMI Performance (CISPR25 Class 5 Peak)

–20
0.1

VOUT
–250V
10mA

C7
0.22µF

FBX

80

–10

R6
47Ω

C9
0.22µF

D3

EN/UVLOSYNC/MODE

0

R8
20Ω

R7
20Ω

D1

SW

LT8365

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
L2: WURTH ELEKTRONIK 74479876147
C3: PANASONIC EEHZC1H680P
C10, C11:12062C104KAT2A
C7, C9: CGJ5L3X7T2D224K160AA
C8: C1206C102KRACTU

D2

D4

1k

CLASS 5 AVERAGE LIMITMEASURED EMISSIONSAMBIENT NOISE

40
30
20
10
0
–10
–20

30

FREQUENCY (MHz)100

1k
FREQUENCY (MHz)12V INPUT TO –250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

8365 TA12c

12V INPUT TO –250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

8365 TA12e

Rev. 0For more information www.analog.com

25

LT8365
TYPICAL APPLICATIONS
Low IQ, Low EMI, 400kHz, 250V Output Boost Converter with SSFM
C10
0.1µF

VIN
12V

D2

INPUT EMI FILTER
L2
470nH
C1
4.7µF
×2
1210
50V

L1
10µH
C2
10µF
50V
1206

C3
68µF
50V

VIN

D1

RT
D1, D2, D3: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324100
L2: WURTH ELEKTRONIK 74479876147
C3: PANASONIC EEHZC1H680P
C10, C11:12062C104KAT2A
C7, C9: CGJ5L3X7T2D224K160AA
C8: C1206C102KRACTU

C70.22µF

FBX

R1
107k

GND

VC
R3
84.5k
C5
1nF

C4
0.22µF

R2
121k

AVERAGE CONDUCTED EMI (dBµV)

PEAK CONDUCTED EMI (dBµV)

50

60
50
40
30
20
10
CLASS 5 PEAK LIMITMEASURED EMISSIONSAMBIENT NOISE
1FREQUENCY (MHz)1012V INPUT TO 250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

40
30
20
10
0
–10
–20

–40
0.1

30
8365 TA13b

10308365 TA13d

Radiated EMI Performance

Radiated EMI (CISPR25
Performance(CISPR25
Class 5 Average)
Class
5 Average)

60

60

50

50

AVERAGE RADIATED EMI (dBµV/m)

PEAK RADIATED EMI (dBµV/m)

1
FREQUENCY (MHz)12V INPUT TO 250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

Radiated EMI Performance

40
30
20
10
0
CLASS 5 PEAK LIMITMEASURED EMISSIONSAMBIENT NOISE100

CLASS 5 AVERAGE LIMITMEASURED EMISSIONSAMBIENT NOISE–30

Radiated EMI (CISPR25
Performance(CISPR25Class
5 Peak)Class 5 Peak)

30

R4
6.49k

C6
1µF

Conducted EMI Performance (CISPR25 Class 5 Average)
60

–20

VOUT

8365 TA13a

70

–10

C8
1nF
500V
1206

INTVCCSS

Conducted EMI Performance (CISPR25 Class 5 Peak)

–20
0.1

VOUT
250V
10mA

BIAS

80

–10

R5
1MΩ

EN/UVLO
LT8365R647Ω

C90.22µF

R720Ω

SW

SYNC/MODE

0

D3

1k

CLASS 5 AVERAGE LIMITMEASURED EMISSIONSAMBIENT NOISE

40
30
20
10
0
–10
–20

30

FREQUENCY (MHz)100

1k
FREQUENCY (MHz)12V INPUT TO 250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

8365 TA13c

12V INPUT TO 250V OUTPUT AT 10mA, fSW = 400kHz, SSFM ON

8365 TA13e

Rev. 026

For more information www.analog.com

LT8365
PACKAGE DESCRIPTION
MSE Package
Variation: MSE16 (12)16-Lead Plastic MSOP with 4 Pins Removed
Exposed Die Pad(Reference LTC DWG # 05-08-1871 Rev D)

BOTTOM VIEW OFEXPOSED PAD OPTION
2.845 ±0.102(.112 ±.004)

5.10(.201)
MIN

2.845 ±0.102(.112 ±.004)

0.889 ±0.127(.035 ±.005)

81

1.651 ±0.102(.065 ±.004)

1.651 ±0.102 3.20 – 3.45(.065 ±.004) (.126 – .136)

160.305 ±0.038(.0120 ±.0015)
TYP

0.50
(.0197)
1.0 BSC(.039)
BSC
RECOMMENDED SOLDER PAD LAYOUT
0.254(.010)

0.35
REF

4.039 ±0.102(.159 ±.004)(NOTE 3)0.12 REF

DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9NO MEASUREMENT PURPOSE

16 14 121110 90.280 ±0.076(.011 ±.003)
REF

DETAIL “A”
0° – 6° TYP

3.00 ±0.102(.118 ±.004)(NOTE 4)4.90 ±0.152(.193 ±.006)

GAUGE PLANE

0.53 ±0.152(.021 ±.006)
DETAIL “A”

1.10(.043)
MAX

0.18(.007)SEATING
PLANE

0.17 – 0.27(.007 – .011)
TYP

1

3 567 81.0(.039)
BSC

0.86(.034)
REF

0.1016 ±0.0508(.004 ±.002)

0.50
MSOP (MSE16(12)) 0213 REV D
NOTE:(.0197)
1. DIMENSIONS IN MILLIMETER/(INCH)
BSC
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALLNOT EXCEED 0.254mm (.010") PER SIDE.

Rev. 0Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license For
is granted
implication orotherwise under any patent or patent rights of Analog Devices.
morebyinformationwww.analog.com

27

LT8365
TYPICAL APPLICATION
400kHz, 3V to 6V Input, 250V Output Boost Converter
C7
0.22µF

D3
VOUT
250V
1.5mA
D2

L1
3.3µH
VIN
3V TO 6V

D1

C1
4.7µF

C6
0.22µF

R8
20Ω

D4

VIN

C5
0.22µFSW
R6
1MΩ

EN/UVLOFBX

LT8365
SYNC/MODE
RT

SS

R3
107k

VOUT

BIAS
INTVCC

GND
C2
0.22µF

VC
R4
84.5k

C4
1µF

R5
6.49k

C3
1nF

D1, D2, D3, D4: DIODES INC. DFLS1150
L1: WURTH ELEKTRONIK 74437324033
C5, C6, C7: CGJ5L3X7T2D224K160AA

8365 TA02a

RELATED PARTS
PART NUMBER

DESCRIPTIONCOMMENTS

LT8300100VIN Micropower Isolated Flyback Converter with
150V/260mA Switch

VIN = 6V to 100V, Low IQ Monolithic No-Opto Flyback, 5-Lead
TSOT‑23

LT833060V, 1A, Low IQ Boost/SEPIC/Inverting 2MHz Converter

VIN = 3V to 40V, VOUT(MAX) = 60V, IQ = 6µA (Burst Mode Operation),

LT8331Low IQ Boost/SEPIC/Flyback/Inverting Converter with
140V/0.5A Switch

VIN = 4.5V to 100V, VOUT(MAX)=140V, IQ = 6µA (Burst Mode
Operation), MSOP-16(12)E

LT8361

100V, 2A, Low IQ Boost/SEPIC/Inverting Converter

VIN = 2.8V to 60V, VOUT(MAX)=100V, IQ = 9µA (Burst Mode
Operation), MSOP-16(12)E

LT8362

60V, 2A, Low IQ Boost/SEPIC/Inverting Converter

VIN = 2.8V to 60V, VOUT(MAX) = 60V, IQ = 9µA (Burst Mode
Operation), MSOP-16(12)E, 3mm × 3mm DFN-10 Packages

LT8364

60V, 4A, Low IQ Boost/SEPIC/Inverting Converter

VIN = 2.8V to 60V, VOUT(MAX) = 60V, IQ = 9µA (Burst Mode
Operation), MSOP-16(12)E, 4mm × 3mm DFN-12 Packages

LT8494

70V, 2A Boost/SEPIC 1.5MHz High Efficiency Step-Up
DC/DC Converter

VIN = 1V to 60V (2.5V to 32V Start-Up), VOUT(MAX) = 70V, IQ = 3µA(Burst Mode Operation), ISD = <1µA, 20-Lead TSSOP

LT85801A (ISW), 65V, 1.5MHz, High Efficiency Step-Up DC/DC
Converter

VIN: 2.55V to 40V, VOUT(MAX) = 65V, IQ = 1.2mA, ISD = <1µA,
3mm × 3mm DFN-8, MSOP-8E

LT395880V, 3.3A Boost/SEPIC/Inverting Flyback Converter

VIN: 5V to 80V, VOUT(MAX) = 83V, IQ = 1.6mA, 5mm × 6mm QFN-36
Package

6-Lead TSOT-23, 3mm × 2mm DFN Packages

Rev. 028

01/20

For more information www.analog.com

www.analog.com

 ANALOG DEVICES, INC. 2020