Buck-Boost PFC Converter with Active Power
Decoupling Capability Using Time-sharing Control
Yuki Miyada
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Niigata, Japans20318786@stn.nagaokaut.ac.jp

Keita Ohata
Dept. of Science of Technology Innovation
Nagaoka University of Technology
Niigata, Japankeita_ohata@stn.nagaokaut.ac.jp

Hiroki Watanabe
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Niigata, Japanhwatanabe@vos.nagaokaut.ac.jp

Jun-ichi Itoh
Dept. of Science of Technology Innovation
Nagaoka University of Technology
Niigata, Japan
itoh@vos.nagaokaut.ac.jp

Abstract—This paper proposes a single-phase buck-boost PFC
converter with active power decoupling (APD) using time-sharing
control. The features of the proposed circuit are; 1) PFC and APD
capability without additional inductors by sharing circuitcomponents and 2) a wide DC voltage control owing to the buckboost operation. The proposed method is characterized by two
operating modes, discontinuous current mode (DCM) andtriangular current mode (TCM). At the DCM method, timesharing control is achieved by operating PFC and APD during
each zero current period. In contrast, the TCM method achieves
zero-voltage switching when switching between PFC and APD
operation. Thus, switching loss is reduced compared to the DCM
method. The validity of the proposed circuit operated in DCM isexperimentally demonstrated by a 200-W prototype. It was
confirmed that the output current ripple at twice the gridfrequency is reduced by 90% in the experiment. Furthermore, theoperation of the proposed circuit operated in TCM isdemonstrated by simulation.
Keywords— single-phase buck-boost PFC converter, active
power decoupling, time-sharing, discontinuous current mode,
triangular current mode, zero voltage switchingI.

INTRODUCTION

Recently, single-phase AC-DC converters are widely
applied in power supplies of home appliances. High harmonics
generation and low power factor by the power supplies cause
problems in power transmission and distribution systems. In
order to solve these problems, a power factor correction (PFC)
circuit has been studied and developed[1]-[3]. Several types of
PFC circuits have been proposed, such as boost, buck, and buckboost types. The buck-boost PFC circuits are used inapplications requiring a wide DC voltage range. In addition, abulky electrolytic capacitor is necessary on the DC link toremove the single-phase power ripple. However, the lifetime ofan electrolytic capacitor is highly dependent on theenvironmental temperature [4]. Moreover, the volume ofelectrolytic capacitors is large, which leads to an increase in thecircuit volume.

In contrast, active power decoupling (APD) circuits havebeen proposed to reduce the size of DC-link capacitors [5]-[19].
The APD circuit uses a small film or ceramic capacitor as abuffer capacitor instead of an electrolytic capacitor. The output
instantaneous power fluctuation is compensated by pulsating thepower of the buffer capacitor at twice the grid frequency.
However, several APD circuits require additional inductors toregulate the buffer capacitor voltage. Therefore, the power
density may decrease due to bulky passive components.
This paper proposes single-phase buck-boost PFC converterwith APD circuit using time-sharing control. The proposed
circuit achieves a wide output voltage range by the four-switch
buck-boost PFC converter. The originality of this paper is thetime-sharing control strategy for APD, buck and boost in one
cycle of switching period. The proposed time-sharing control
has no control interference among APD, buck and boost.
Furthermore, the number of inductors and switching devices are
reduced by sharing circuit components.
In this paper, two types of inductor current control modes are
explained to achieve time-sharing control. First, the proposed
circuit's operating method in discontinuous current mode
(DCM) is discussed. The validity of the proposed circuit
operated in DCM is experimentally demonstrated by a 200-W
prototype. Next, the method of operating in triangular current
mode (TCM) is discussed. The operation of the proposed circuit
operated in TCM is demonstrated by simulation.
II. PROPOSED PFC CONVERTER CONFIGURATION
Fig.1 shows the proposed single-phase buck-boost PFC
circuit with an APD circuit. The APD circuit consists of only asmall buffer capacitor and the switching device of SW3. The
buck-boost PFC converter is used for the PFC capability, DC
voltage control, and active power decoupling capability. The ACside leg of the buck-boost PFC circuit is Leg 1 and the DC side
leg is Leg 2.

Active power decoupling circuit
Iin Lf

SW3 SW1p

Vin

SW2p
IL

100V

Cf

Vdc

ICbuf

SW1n
Leg1

PFC

APD

iL
0

Vout

SW2n

Leg2

Fig. 1. proposed circuit configuration.

SW1p
SW1n
SW2p
SW2n
SW3

SW1p
SW1n
SW2p
SW2n
SW3
T1

T2

T3

T1

T4

T2

Tsw

Vdc > Vout : Leg2 driving mode (boost operation)3.

Vdc ≈ Vout : 4-arm driving modeMoreover, the converter has two modes for APD operation:1.

Cbuf charge mode2.

Cbuf discharge mode

Fig. 2, Fig. 3 and Fig. 4 show the inductor current waveformand switching pattern for one switching period. In the proposed
circuit, PFC and APD operation is achieved within one
switching period by using time-sharing control by DCM. Thus,
there are six control patterns in one switching period. Fig. 2
shows the waveforms of the Leg1 driving mode, Fig. 3 shows
the Leg2 driving mode, and Fig. 4 shows the 4-arm driving mode.
The current of Cbuf flows through the inductor L from the output
side by switching SW1p and SW1n during the zero current periodof PFC operation.
Generally, a buck-boost operation for PFC is implemented
using Leg1 driving and Leg2 driving mode. However, the input
current is distorted due to an over-modulation in the condition
that Vdc and Vout are almost same. In order to solve this problem,the 4-arm driving mode is added. In 4-arm driving mode, SW1p
and SW2n switches in period T1, and SW1n and SW2p switches inperiod T2. The inductor voltage VL does not be small in thecondition that Vdc and Vout are almost same, thus overmodulation does not occur in 4-arm driving mode. Hence, theproposed circuit achieves low harmonic distortion current
waveforms. During the PFC operation, the APD circuit is
disconnected from the PFC circuit by turning SW3 off, andduring the APD operation, SW3 is turned on to charge anddischarge the Cbuf. The buffer capacitor voltage VCbuf iscontrolled to be higher than the peak of input voltage and Vout.
Thus, the body diode of SW3 does not conduct.
Fig. 5 shows the input current control block diagram. The
duty ratios of periods T1 ~ T4 in Fig. 2, Fig. 3 and Fig. 4 aredefined as d1 ~ d4. These duties are calculated from the inductor
current in DCM condition. The duty ratios in each mode are
given by the following equations:

T4(b) discharge mode

Fig. 2. Inductor current waveform and switching pattern for one
switching period at Leg1 driving mode.

PFC

APD

APD
discharge

PFC

iL

Vdc < Vout : Leg1 driving mode (buck operation)

2.

T3
Tsw

(a) charge mode

III. CONTROL METHOD FOR DCM OPERATION
The proposed PFC converter has three modes for the Voutcontrol depending on the relationship between the diode rectifier
output voltage Vdc and the output voltage Vout as follows;
1.

APDdischargeiL
charge

0

VL
L

VCbuf

Cbuf

PFC

Iout

iL
charge

0
SW1p
SW1n
SW2p
SW2n
SW3

0
SW1p
SW1n
SW2p
SW2n
SW3

T1

T2

T3

T4

T1

T2

Tsw

T3

T4

Tsw

(a) charge mode(b) discharge mode

Fig. 3. Inductor current waveform and switching pattern for one
switching period at Leg2 driving mode.

PFC

APD

PFC

iL

APD
dischargeiL
charge

0
SW1p
SW1n
SW2p
SW2n
SW3

0
SW1p
SW1n
SW2p
SW2n
SW3

T1

T2

T3

T4

Tsw
(a) charge mode

T1

T2

T3

T4

Tsw
(b) discharge mode

Fig. 4. Inductor current waveform and switching pattern for one
switching period at 4-arm driving mode.

<For Leg1 driving mode>
d1 _ Leg1 =2Liin* ............................... (1),
(Vdc − Vout )Tsw

Vdc − Vout................................... (2),Vout
<For Leg2 driving mode>d 2 _ Leg1 = d1 _ Leg1

d1 _ Leg 2 =2 L(Vout − Vdc ) *iin ............................... (3),
VdcVout Tsw

2L(Vdc - Vout)Tsw

Vdc.................................. (4),Vout − Vdc<For 4-arm driving mode>d 2 _ Leg 2 = d1 _ Leg 2

d1 _ 4 − arm =Vdc + Vout............................... (6),
Vout

where iin* is the input current command, Vdc is the diode rectifier
output voltage, Vout is the output voltage, L is the inductance, and
Tsw is the switching period. These duties are uniquely determinedfrom (1), (2), (3), (4), (5), and (6). Hence, any feedback control
is not needed for input current control.
Fig. 6 shows the control block diagram for buffer capacitor
voltage control. The input instantaneous power pin under theunity power factor is given bypin = Vm I m sin 2 ( grid t )................................. (7),Vm I m1 − cos(2 grid t )2
where Vm and Im are the maximum values of grid voltage andcurrent, and ωgrid is the grid angular frequency. The second termin (1) means that the input instantaneous power fluctuates attwice the grid frequency. Thus, the buffer capacitor voltage VCbuf
must be controlled to fluctuate with the frequency of 2ωgrid. The
inductor current command is given by=iLbuf * =Pncos(2grid t ) ....................................... (8),Vout

where Pn is the input power. A wide bandwidth control in the PI
controller for VCbuf is not required because the average voltageof buffer capacitor VCbuf_ave and the command value VCbuf* are notinstantaneous values, respectively. Note that the switching
between charging and discharging operations is determined by
the positive or negative polarity of the inductor current
command value iL*.
IV. EXPERIMENTAL RESULT AT DCM OPERATION
Table. 1. shows the experimental conditions. The operationof the prototype circuit was tested under the following two
voltage and power conditions.
Buck-boost operation condition :2L(Vout - Vdc )Vdc VoutTsw

2L
Vdc Tsw

2L
iin* ........................................ (5),
Vdc Tsw

d 2 _ 4 − arm = d1 _ 4 − arm

1.

iin*

sqrt

d1_Leg1
Vdc - Vout
Vout

sqrt

d1_Leg2
Vdc
Vout - Vdc

sqrt

VCbuf

LPF

+
-

PI

ICbuf_ave *

VCbuf
Vout

d2_Leg2
d1_4-arm

Vdc +Vout
Vout

d2_4-arm

Fig. 5. Proposed input current control block for buck-boost PFC
converter.
Table. 1. Experimental parameters at DCM operation

Items

Value

Grid voltage Vin
Grid angular frequencyωgrid
Inductor L
Buffer capacitor Cbuf
Input filter inductor Lf
Input filter capacitor Cf
Switching frequency fsw
Voltage control natural frequency ωnVoltage control damping factor ζ

100 Vrms
2π ∙ 50 rad/s
33 μH
47 μF
1.3 mH
1 μF
50 kHz
100 rad/s
0.707

2.

Boost operation condition : Vout = 200 V, Pn = 200 W

Fig. 7 (a) shows the operating waveforms under the buckboost operation condition (Vout = 100V, Pn = 100W). The buffer
voltage command VCbuf* is set to 230V. In the buck-boost
operation condition, the operation mode switches between Leg1,
Leg2, and 4-arm driving modes within the grid period by the
relationship of the voltage amplitude between Vdc and Vout. The
4-arm driving mode is applied when the condition of | Vdc - Vout
| < 20V is satisfied. Fig. 7 (a) shows that the proposed circuitachieves an input current power factor of 0.98 and an input
current THD of 5.2%. In addition, the efficiency was 89.5%
under the buck-boost operation condition. Fig. 7 (b) shows theoperating waveforms under the boost operation condition (Vout= 200V, Pn = 200W). Active power decoupling was applied, andthe average value of VCbuf was set to 300V. Fig. 7 (b) shows thatthe proposed circuit achieves an input current power factor of
0.99 and an input current THD of 5.3%. In addition, theefficiency was 90.2% under the boost operation condition.
Fig. 8 (a) shows the output current harmonics analysis
results for the buck-boost operation condition, and Fig. 8 (b)shows the boost operation condition. Fig. 8. shows that the 2ndorder harmonics (100 Hz) of the output current are reduced bymore than 90% by applying APD under both conditions.

Vout = 100 V, Pn = 100 W

VCbuf_ave *d2_Leg1

Pn cos(2ωgridt)
Vout
iLbuf*
iL_ave*iL* 2L(VCbuf - Vout)+
VCbufVoutTsw
+

sqrt

BEF
Fig. 6. Proposed buffer capacitor voltage control block diagram for APD circuit.

d3
Vout
VCbuf - Vout

d4

Leg2

4-arm
Leg1

Leg2

Vin, Vout, VCbuf [200V/div]

Vin, Vout, VCbuf [200V/div]
0V

0V
Input current Iin [2A/div]
THD:5.16%

0A

Iin [2A/div]
THD:5.28%

0A

Inductor current IL [10A/div]
0A

IL [10A/div]
0A[4ms/div][4ms/div](a) buck-boost operation condition(b) Boost operation condition(Vout = 100V, Pn = 100W)(Vout = 200V, Pn =200W)
Fig. 7. Operating wa veform when the proposed DCM -based time-sharing control is applied.

:0.68A)

8080
70

50

2020
10

90% reduces

22(100Hz)

44

6
6

88

10
10

CONTROL METHOD FOR TCM OPERATIONIn this chapter, the time-sharing control using TCM is
explained. This control method extends the time-sharing control
using DCM described in the previous chapter. By operated in
TCM, it is possible to achieve ZVS[20]-[22] when operation isswitched, and high efficiency is achieved through the reductionof switching losses.
Fig. 9, Fig. 10 and Fig. 11 show the inductor current
waveform and switching pattern for one switching period at
TCM. The PFC operation is applied in the first half of one
switching period, and the APD operation is applied in the second
half. Depending on the voltage conditions, one of three modes
for the PFC operation is selected from Leg1, Leg2, and 4-arm
driving modes. In addition, one of two modes for the APD
operation from Cbuf charging and discharging mode. Hence,
there are six control patterns in one switching period, the sameas in the DCM method. Fig. 9 shows the waveforms of the Leg1
driving mode, Fig. 10 shows the Leg2 driving mode, and Fig. 11
shows the 4-arm driving mode. In the proposed method, TCM

With active power
decoupling

2020
10

00
66
8 10
12
16
18
12
18
0
44
14
14 16
20
12
14
16
18
20
0
22
8
10
12
14
16
18
(100Hz)Harmonic number
Harmonic number(a) buck-boost operation condition(b) Boost operation condition(Vout = 100V, Pn = 100W)(Vout = 200V, Pn =200W)
Fig. 8. Output current harmonics analysis results when the proposed DCM -based time-sharing control is applied .

0
0

200W

Without active power
decoupling

(100%

30

From the experimental results, the proposed circuit andcontrol method has been achieved for PFC and APD operation.
However, the problem of the proposed method is that the
maximum efficiency is only 90.2%. One of the reasons for thisproblem is that zero voltage switching (ZVS) is not achieved at
DCM operation when the operation is switched between PFC
and APD.
V.

8080

4040 power
With active
decoupling
30

4040

00

Without active power decouplingWith active power decoupling90

70 active power
Without
decoupling
6060

97% reduces

50

100
100W

: 0.72A: 0.67A

6060

100:0.71A)

Without active power decouplingWith active power decoupling(100%

Harmonic component ratio
based on DC component[%]

100

90

Harmonic component ratio
based on DC component[%]

100

20
20

operation is achieved by the negative current flow during theswitching time from PFC to APD operation (T2 - T3 period), as
shown in Fig. 9, Fig. 10 and Fig. 11. The inductor current is
controlled separately for PFC and APD operation by switching
the operation with the set bottom current Ibot and top current Itop.
TCM operation achieves ZVS by the negative current ofswitching devices during the dead time. Thus, the switching
period Tsw must be controlled in TCM according to the current
command value in order to achieve ZVS. In addition, Tsw isrelated to achieving two current command values, which are
PFC and APD operation. Hence, the Tsw is determined by threecurrent commands which are for the required negative current ofswitching devices for ZVS, the PFC current and the APD current.
Fig. 12 shows the control block diagram of the proposed
control method. The input current command iin* is calculated
based on the power command value. The APD current commandiAPD* is the same as in the DCM method: the charge/discharge
current command iLbuf*(8) is added to the output of the PI
controller that controls the average buffer capacitor voltage
VCbuf_ave. The power pulsation is compensated by pulsating VCbuf
through adding iLbuf* to the PI controller output.
The operating time of each switching pattern is expressed as
T1, T2, T3 and T4. The PFC operation period T1, the APD
operation period T3 and switching period Tsw must be calculated
from the operating conditions. In contrast, T2 and T4 are uniquelydetermined from the time to reach the bottom current Ibot and top
current Itop, respectively. Thus, the derivation of Tsw, T1, and T3

is explained using the Leg 1 driving mode and discharge mode.
Tsw is derived so that the current average current values of PFC
and APD operations within one switching period are equal toeach current command value. However, deriving Tsw requires
solving nonlinear simultaneous equations that need to beobtained by numerical analysis. The switching frequency Tsw is
not uniquely derived from an analytical calculation. Therefore,
in this paper, Tsw is derived by approximating the equation as Ibot
= Itop = 0, which makes the equation analyzable. Tsw for Leg 1
drive and discharge mode, which is derived using approximation,is expressed in (9).

V V 2 i* i* V
− Vout ) 4L V i*  Cbuf dc APD in ( Cbuf
Tsw =Cbuf APDVout (Vdc − Vout )Vout (VCbuf − Vout ) 


*
2 *
2 L (Vdc2Vout iin* − VCbuf Vdc2 iin* + VCbuf VdcVout i APD− VCbuf Vouti APD )−

2
Vout
(VCbuf − Vout ) (Vdc − Vout ) VCbuf+
2 Li VCbuf − Vout  bot.......................................................................... (9).

The command values for periods T1 and T3 are derived basedon (9). The period T1 is derived from the input current commandand the period T3 is derived from the APD current command
value using (10) and (11).
2 LTsw
L
iave _ PFC +ibot ................... (10)Vdc − Vout
VCbuf − Vout

T1 =2
2 LVoutTsw
2 L2ibotiave _ APD +VCbuf (VCbuf − Vout )(VCbuf − Vout ) 2

T3 =
+

L
ibot
VCbuf − Vout

APD
charge

0

PFC
iL

Itop
Ibot

0

SW1p

SW1p

SW1n

SW1n

SW2p

SW2p

SW2n

SW2n

SW3

SW3
T2

T1

T3

APD
discharge
Ibot

T2

T1

T4

(a) charge mode

SIMULATION RESULT AT TCM OPERATION

The proposed control method is verified by simulation under
two voltage and power conditions similar to the experimental
conditions at DCM operation. Table. 2. shows the simulation
conditions. The bottom current Ibot and top current Itop were set
to 0.5 A for both conditions. The output current Iout_LPF is the
waveform passed through an LPF for observation with a 1 kHz
cutoff frequency.
Fig. 13 (a) shows the operating waveforms under the buckboost operation condition (Vout = 100V, Pn = 100W). The output
current Iout_LPF is the waveform passed through an LPF for
observation with a 1 kHz cutoff frequency. The active power
decoupling was applied, and the average value of VCbuf was set
to 230V. In the buck-boost operation condition, the operation
mode switches between Leg1, Leg2, and 4-arm driving modes
within the grid period by the relationship of the voltage
amplitude between Vdc and Vout. Fig. 13 (a) shows that the
proposed circuit achieves an input current power factor of 0.99

T4(b) discharge mode

Fig. 9. Inductor current waveform and switching pattern for one
switching period at Leg1 driving mode

PFC
iL

APD
charge

0

PFC
iL

Itop
Ibot

APD
discharge
Itop
Ibot

0

SW1p

SW1p

SW1n

SW1n

SW2p

SW2p

SW2n

SW2n
SW3

SW3
T1

T2

T3

T1

T4

T2

T2-2

T3

T4

Tsw

(a) charge mode....... (11)

T3

Tsw

Tsw

Tsw

Note that the results of deriving Tsw, T1, and T3 for the otheroperating modes are shown in the Appendix.
VI.

PFC
iL

(b) discharge mode

Fig. 10. Inductor current waveform and switching pattern for one
switching period at Leg2 driving mode

PFC
iL

APD
charge

0

Itop
Ibot

PFC
iL
0

SW1p

SW1p

SW1n

SW1n

SW2p

SW2p

SW2n

SW2n

SW3

SW3
T1

T2

T3

Tsw
(a) charge mode

T4

APDdischarge
Ibot

T1

T2

T3

T4

Tsw
(b) discharge mode

Fig. 11. Inductor current waveform and switching pattern for one
switching period at 4-arm driving mode

and an input current THD of 3.2%. Fig. 9 (b) shows theoperating waveforms under the boost operation condition (Vout= 200V, Pn = 200W). The active power decoupling was applied,
and the average value of VCbuf was set to 300V. Fig. 13 (b) showsthat the proposed circuit achieves an input current power factorof 0.98 and an input current THD of 4.7%.
Fig. 14 shows the results of the harmonic analysis of theoutput current. Fig. 14 shows that the 2nd-order harmonics (100Hz) of the output current are reduced by more than 95% in bothconditions compared to without APD.

Pn
Vin

2 sin(ωgridt)

Vin
Vout
VCbuf

VII. CONCLUSION
This paper proposed a single-phase buck-boost PFC
converter with active power decoupling using the time-sharing
control. The proposed circuit achieves a wide output voltage
range. In addition, the time-sharing control achieves high power
density by sharing the inductor and the devices between PFC
and APD operation. The time-sharing control using two typesof inductor current modes, DCM and TCM, are proposed. The
DCM method was tested under two conditions: buck-boost andboost operation conditions. From the experimental results, it
was verified that the input current THD was 5.3% and thepower factor was 0.98 under both conditions. In addition, the2nd-order harmonics (100 Hz) of the output current was
reduced by 90%. However, the DCM method has the problem
that its maximum efficiency is only about 90%. In contrast, theTCM method reduces switching loss compared to the DCM
method by ZVS. The simulation results of the TCM method
showed that the proposed control method achieves fundamental

VCbuf_ave *
VCbuf

+

LPF

-

PI

ICbuf_ave * VCb u f +
Vout

Vin
Vout
VCbuf

iin*

*

Cal. of

Tsw*

iAPD *

Tsw*
Vout
VCbuf

Cal. of

T1*

Cal. of

+

T4*

4-arm

4-arm

VCbuf

Fig. 12. Control block diagram at TCM operation.
Table 2. Experimental parameters at TCM operation

Items

Value

Grid voltage Vin
Grid angular frequencyωgrid
Inductor L
Buffer capacitor Cbuf
Input filter inductor Lf
Input filter capacitor Cf
Voltage control natural frequency ωnVoltage control damping factor ζ

100 Vrms
2π ∙ 50 rad/s
250 μH47 μF
1.3 mH
5 μF
100 rad/s
0.707

PFC and APD operation.
In future works, the efficiency of the TCM method will becompared with that of the DCM method based on theexperimental results.

[200V/div][V] 400

Vin
Vout[200V/div]

VCbuf

Vout

200

0
-200
2.0

0.0

Vin
0

Iin [2A/div]
THD:4.7%
PF:0.98-200[A] 4.0

Iin[2A/div]
PF:0.99
THD:3.2%

2.0
0.0

5.0

-2.0
-4.0
[A] 10

0.0

5

-2.0

-5.0
1.5

0.0

IL [4A/div]-5.0

Iout_LPF [0.5A/div]

IL[5A/div][A] 1.5

Iout_LPF[0.5A/div]

1.0

1.0

0.5
0.5

[10ms/div]

0.0
0.0

[10ms/div]0.0(a) buck-boost operation condition(Vout = 100V, Pn = 100W)T3*

Pn *cos(2ωgridt)
iL_buf*
Vout

BEF

Operation mode
Leg2
Leg1
Leg2

200

T1*

(b) boost operation condition(Vout = 200V, Pn =200W)

Fig. 13. Simulation results when the proposed TCM-based time-sharing control is applied.

100

100
100W

:0.98A)80
80

Harmonic component ratio
based on DC component[%]

Without active power decouplingWith active power decoupling

90

95% reduces

50

60

50

Without active power
decoupling

97% reduces

40
40 power
With active
decoupling
30

30
20
20
10

00

2

2
(100Hz)

66

88
10
12
16
14
10
12
14
16
Harmonic number(a) buck-boost operation condition(Vout = 100V, Pn = 100W)
44

18
18

20
20
10
00

20
20

With active power
decoupling

(100%

40
40

00

80
80

: 0.96A: 0.96A

60
60

200W

Without active power decouplingWith active power decoupling

90

70active power
Without
decoupling
60

70

(100%

Harmonic component ratio
based on DC component[%]100:0.98A)

100

22

00

(100Hz)

66

88
10
12
14
10
12
14
Harmonic number(b) boost operation condition(Vout = 200V, Pn =200W)
44

16
16

18
18

20
20

Fig. 14. Output current harmonics analysis results when the proposed TCM -based time-sharing control is applied .

APPENDIX

2 L (VCbuf − Vout ) Tsw

T3 =VoutVCbuf

Tsw, T1, and T3 in Leg 1 driving and charge mode are derived
as follows ;2 LTsw
L
iave _ PFC +ibot
Vin − Vout
VCbuf − Vout

T1 =

2 L (VCbuf − Vout ) Tsw

T3 =VoutVCbuf

iave _ APD +

Li

2
Vout
VCbuf

............................................................................... (A-8)
+

L
itop
Vout

...... (A-2)+(

2 L −V V i+ VCbuf V i

2
in ave _ PFC

2
out

V

(V

Cbuf

− VCbuf VinVout iave _ APD + VCbuf V2out− Vout ) (Vin − Vout )iave _ APD)....................................................................... (A-3)

Tsw, T1, and T3 in Leg 2 driving and discharge mode arederived as follows ;*22 L(Vout − Vin )iPFC
Tsw − L2itop

T1 =VinVout

+

L
VCbuf

............ (A-4)
ibot

(

2
2
L2ibot
VCbuf+ VoutVCbuf − Vout2
2 LVoutTsw
T3 =i*APD +
VCbuf (VCbuf − Vout )VCbuf 2 (VCbuf − Vout ) 2VCbuf Viniave _ APDiave _ PFC (Vout − Vin ) 4LVout iave _ PFC +
Vin (Vout − Vin ) VCbuf − Vout




3
2
2 L (VCbuf Vin2iave _ APD − VCbuf VinVout iave _ APD − Voutiave _ PFC + VCbuf Voutiave _ PFC )Tsw, T1, and T3 in Leg 2 driving and charge mode are derived
as follows ;T1 =2
L2itop2 L(Vout − Vin )Tsw *L
iPFC ++
ibot
VinVout
VinVout VCbuf......... (A-7))

Tsw, T1, and T3 in 4-arm driving and discharge mode arederived as follows ;................................... (A-10)

T1 =2 LTsw *L
iPFC +
ibot
Vin
VCbuf

T3 =

2
2
2
L2ibot− Vout( 2VCbuf)
2 LVoutTsw
i*APD + 2
2
VCbuf (VCbuf − Vout )VCbuf (VCbuf − Vout )

+

L
ibot
VCbuf − Vout..................................................................... (A-11)

VCbuf iave _ APDiave _ PFC (VCbuf − Vout ) V iCbuf ave _ APD + (Vin + Vout )VinVout
Vout (VCbuf − Vout ) 


3
2
2
2
2 L(Vout iave _ PFC − VinVCbuf iave _ PFC − VoutVCbuf iave _ PFC + VinVout iave _ PFC

−

4L

2
VinVout(VCbuf − Vout )+Vin2Vout iave _ PFC + VCbuf VinVout iave _ APD − 2VCbuf VinVout iave _ PFC )VinVout (VCbuf − Vout ) (Vin − Vout )....................................................................... (A-6)

VinVout (VCbuf − Vout ) (Vin − Vout )....................................................................... (A-9)

Tsw =....................................................................... (A-5)

+

(

3
2
2 L −VCbuf Vin2 iave _ APD + VCbuf VinVout iave _ APD + Voutiave _ PFC − VCbuf Voutiave _ PFC

)

L
+
ibot
VCbuf − Vout
Tsw =

+

2

Vout
VCbuf (VCbuf − Vout )iave _ PFC iave _ APD VCbuf iave _ APD +
Vin (Vout − Vin )Vout (VCbuf − Vout ) 



4L

Tsw =


VCbuf Vin2 (VCbuf − Vout )iave _ PFC iave _ APD 
4L
V

Tsw =
Cbuf iave _ APD +
Vout (Vin − Vout )Vout (VCbuf − Vout )2in out ave _ PFC

222
L2 (VCbuf − Vout ) (VCbuf ibot
− Vout itop)
L2ibot+2
2
VoutVCbuf
VoutVCbuf

L
+
itop
Vout

.................... (A-1)

2 2
top

i*APD +..................................................................... (A-12)

Tsw, T1, and T3 in 4-arm driving and charge mode are derived
as follows ;T1 =2 LTsw *L
iPFC +
ibot
Vin
VCbuf

................................... (A-13)

222 L (VCbuf − Vout ) Tsw *L2ibot(VCbuf − Vout ) + L2itop2 (VCbuf − Vout )L2 ibot
iAPD +−
2
2
VoutVCbuf
VoutVCbuf
VoutVCbuf
Vout
VCbuf

T3 =
+

L
itop
Vout

............................................................................ (A-14)

VCbuf (VCbuf − Vout )iave _ PFC iave _ APD VCbuf iave _ APD + (Vin + Vout )VinVout
Vout (VCbuf − Vout ) 


3
2
2
2
2
2 L(Vout iave _ PFC − VCbuf Vin iave _ PFC − VCbuf Vout iave _ PFC + 2VinVout iave _ PFC + VinVout iave _ PFC

Tsw =
−

4L

2
VinVout(VCbuf − Vout )−2VCbuf VinVout iave _ PFC + VCbuf VinVout iave _ APD )..................................................................... (A-15)

REFERENCES
[1]

[2]

[3]

[4]

[5]

[6]

[7]

[8]

Z. Bing, M. Chen, S. Miller, Y. Nishida, and J. Sun, “Recent
developments in single-phase power factor correction,” inProc.Power
Conversion Conference-Nagoya, 2007. PCC’07, 2007, pp.1520–1526.
L. Huber, L. Gang, and M. M. Jovanovic, “Design-oriented analysisand
performance evaluation of buck PFC front end,”IEEE Trans.PowerElectron., vol. 25, no. 1, pp. 85–94, 2010.
G. Andersen, “Current programmed control of a single-phase two-switch
buck-boost power factor correction circuit,”IEEE Trans. Ind.Elec., vol.
53, no. 1, pp. 263–271, 2006.
H. Hu, S. Harb, X. Fang, D. Zhang, Q. Zhang, Z. J. haen: "A Three-port
Fly back for PV Micro inverter Applications With Power Pulsation
Decoupling Capability", IEEE Trans., Vol. 27 No. 9, pp. 3953-3964(2012)
D. Neumayr, G. C. Knabben, E. Varescon, D. Bortis and J. W. Kolar,“Comparative Evaluation of a Full- and Partial-Power Processing Active
Power Buffer for Ultracompact Single-Phase DC/AC Converter Systems”,
IEEE Journal of Emerging and Selected Topics in Power Electronics, Vol.
9, No. 2, pp. 1994-2013, 2021.
Y. Cui, H. Han, Y. Liu, G. Xu, M. Su, and S. Xie, “An EfficiencyImproved Single-Phase PFC Rectifier With Active Power Decoupling”,
IEEE Transactions on Power Electronics, Vol. 37, No. 9, pp. 1078410796, September, 2022.
C. B. Barth, I. Moon, Y. Lei, S. Qin, R. C. N. Pilawa-Podgurski:“Experimental Evaluation of Capacitors for Power Buffering inSinglePhase Power Converters”, ECCE 2015, pp. 6269-6276 (2015)
S. Yamaguchi, T. Shimizu : "Single-phase Power Conditioner with a
Buck-boost-type Power Decoupling Circuit", IEEJ Journal of Industry
Applications, Vol. 5, No. 3, pp. 191-198 (2016)[9]

Montie. A, Luciano. F.S Alves, Ruxi. Wang, and Mauricio. B.R. Correa,“Low-Frequency Power Decoupling in Single-Phase Applications
overview,”IEEE Transactions on Power Electronics, vol: pp. (2016)[10] Y. Tang, F. Blaabjerg, “An efficiency improved active power decoupling
circuit with minimized implementation cost” 2014 international powerelectronics and application conference and exposition. pp. 864-869(2014)[11] S. Kjaer and J. Pedersen, “A review of single-phase grid-connected
inverters for photo-voltaic modules,” IEEE Trans. Ind. Ap., vol. 41, no. 5,
pp. 1292-1306, (2005)[12] S. Ma, H. Wang, H. Tang, G. Zhu and H. Wang “Lifetime estimation of
DC-link capacitors in a single-phase converter with an integrated active
power decoupling module” IECON 2016, pp. 6824-6829 (2016)[13] V.V.S. Pradeeo Kumar, B.G. Fenabdes “transformerless active power
decoupling topologies for grid connected PV applications” IECON 2016,
pp. 2410-2419 (2016)[14] I. Serban, C. Marinescu, D. Munteanu “Performance analysis of a SiCbased single-phase H-bridge inverter with active power decoupling” EPE
2016 pp.1-10 (2016)[15] S. Qin, Y. Lei, C. Barth, W. -C. Liu and R. C. N. Pilawa-Podgurski, “A
high-efficiency high energy density buffer architecture for power
pulsation decoupling in grid-interfaced converters,” 2015 IEEE Energy
Conversion Congress and Exposition (ECCE), pp. 149-157, 2015.
[16] Y. Sun, Y. Liu, X. Li, J. Yang “Active power decoupling method forsingle-phase current source rectifier with no additional active switches”
IEEE trans. vol. 31, pp.5644-5654 (2016)[17] N. Pragallapati, V. Agarwal, “Single phase solar PV module integrated
flyback based micro-inverter with novel active power decoupling” PEMD
2014. pp. (2014)[18] W. Qi, H. Wang, X. Tan, G. Wang, K. D.T Ngo, “A novel active power
decoupling single-phase PWM rectifier topology” APEC 2014. pp. 89-95(2014)[19] T. Hirao, T. Shimizu, M. Ishikawa, K. Yasui “Discussion on Modulation
Methods for Flyback-type Single-phase Inverters with Enhanced Power
Decoupling for Photovoltaic AC module Systems” IEEJ trans., No.4, pp.
504-510 (2006)[20] J. Zhang, J-S. Lai, R-Y. Kim, W. Yu, “High-Power Density Design of aSoft-Switching High-Power Bidirectional dc–dc Converter”, IEEE
Transaction on Power Electronics, Vol. 22, No. 4, pp. 1145 - 1153 (2007).
[21] C. Marxgut, F. Krismer, D. Bortis, J. W. Kolar, “Ultraflat Interleaved
Triangular Current Mode (TCM) SinglePhase PFC Rectifier”, IEEE
Transaction on Power Electronics, Vol. 29, No. 2, pp. 873 - 882 (2014).
[22] X. Huang, F. C. Lee, Q. Li, W. Du, “HighFrequency High-Efficiency
GaN-Based Interleaved CRM Bidirectional Buck/Boost Converter with
Inverse Coupled Inductor”, IEEE Transaction on Power Electronics, Vol.
31, No. 6, pp. 4343 - 4352 (2015).