Adjustable Field PM Motor Based on Permeability
Modulation Technique Using Zero-Phase Current
Toshihiko Noguchi
Graduate School of Integrated Scienceand Technology
Shizuoka University
Hamamatsu, Japan
noguchi.toshihiko@shizuoka.ac.jp

Kiyohiro Iwama
Graduate School of Integrated Scienceand Technology
Shizuoka University
Hamamatsu, Japan
iwama.kiyohiro.14@shizuoka.ac.jp

Masahiro Aoyama
Graduate School of Integrated Scienceand Technology
Shizuoka University
Hamamatsu, Japan
aoyama.masahiro@shizuoka.ac.jp

Abstract— An adjustable field magnetic flux permanent
magnet (PM) motor is proposed in this paper, which is based on apermeability modulation technique. The proposed PM motor hasmagnetic paths made of soft magnetic metal between thepermanent magnet poles in the rotor, and a special coil for thepermeability modulation of the soft magnetic metal. The
proposed PM motor can control the amount of the leakage flux
between the rotor permanent magnet poles by means of magnetic
saturation of the soft magnetic metal, where the main magnetic
flux interlinking to the stator windings is effectively reduced
because the leakage flux between the permanent magnet poles isdominant.
However, when the soft magnetic metal ismagnetically saturated by the zero-phase magnetic flux, theleakage magnetic flux is reduced, resulting in the enhancement of
the main magnetic flux to the stator. This paper demonstrates
computer simulation results of the proposed strategy for the
adjustable field magnetic flux PM motor.

control the magnetic flux freely particularly in a low speed
range. Furthermore, almost all the adjustable magnetic flux
PM motors are hard to generate reluctance torque, so they havean essential demerit of low power density, compared withstandard interior permanent magnet (IPM) motors.

Keywords—3D magnetic path, permeability modulation,
adjustable field, PM motor, magnetic saturation, zero-phase current

A. Adjustable Magnetic Flux Technique of Principle Model
Fig. 1 shows a basic principle of the permeability
modulation used for adjusting the magnetic flux. As can beseen in the figure, the main magnetic flux is horizontallypenetrating the soft magnetic metal, which is adjusted by thedifferent magnetic flux for permeability modulation penetrating
the soft magnetic metal vertically. By intensifying themagnetic flux for the permeability modulation, the soft
magnetic metal is magnetically saturated, and its permeability
is lowered, resulting in the decrease of the main magnetic flux
horizontally flowing into the soft magnetic metal. Therefore, it
is possible to control the main magnetic flux amount by using
the different magnetic flux for the permeability modulation.

I. INTRODUCTION
Permanent magnet (PM) motors have widely been used forvarieties of applications such as automotive and industry
equipment. The PM motors have achieved higher power
density than any other different types of motors, taking great
advantage of high-energy density PM such as NdFeB. It isdifficult, however, to design a particular PM motor for lowspeed-high-torque applications and high-speed-low-torqueapplications at the same time because of the constant magnetic
field caused by the PM. Therefore, a field weakening
technique has been employed by a negative injecting d-axis
current to the motor, but the significant copper loss increase
due to the d-axis current injection can cause serious
degradation of the efficiency.
In order to solve the problems described above, adjustable
magnetic field PM motors are intensively discussed in recent
years [1]-[5]. The refferences [1]-[4] describe techniques toadjustable operation of an air gap flux by using static magnetic
field generated by a field winding. However, there are someproblems about power losses of a DC/DC converter for thefield windings, and other problems on higher power losses by afield winding also remain. On the other hand, the reference [5]reports a different adjustable magnetic flux technique using
inverse magnetic poles generated by a second-order spaceharmonic component. The technique makes it possible to978-1-7281-3398-0/19/$31.00 ©2019 Crown

In this paper, a novel technique to achieve
electromagnetically the adjustable magnetic flux PM motor ispresented, which has inverse saliency to generate the reluctancetorque as well as the PM torque. Some computer simulation
results are demonstrated to show the performance of the
proposed strategy, followed by the basic operation principle
and a magnetic circuit design method of the PM motor.
II. ADJUSTABLE MAGNETIC FLUX TECHNIQUE BASED ON
PERMEABILITY MODULATION

Fig. 2 illustrates an application of the permeability
Magnetic flux for permeability modulation
Permanent magnet magnetic flux

Soft magnetic metal(a) Without magnetic saturation.

(b) With magnetic saturation.

Fig. 1. Permeability modulation principle.

B. Feasibility Study of Proposed Adjustable

Leakage magnetic flux

Rotor

Soft magnetic metal(a) Without magnetic saturation.

(b) With magnetic saturation.

Fig. 2. Principle of proposed motor.

Ferrite magnet(Br = 0.4 T)

StatorPermeability
modulation coil

Magnetic steel sheet
z

U

Rotor

x
StatorRotoryxSoft magnetic material(MB1H)(a) Cross section.

Soft magnetic composite(b) Axial cross section.

Fig. 3. Principle motor model.

Soft magnetic metal

Magnetic Flux Operation

Magnetic field
(A/m)
5000

Figure 3 shows a principle model of the adjustable
magnetic flux IPM motor, which has a 4-pole and concentrated
winding configuration. As can be seen in Fig. 3(b), the special
winding for the permeability modulation is placed at the bothends of the stator so as to have a z-axis direction magnetic flux
path. All the following electromagnetic analyses are conductedwith JMAG-Designer 17.0 TM.
Figure 4 shows magnetic field intensity distribution of thesoft magnetic metal when the magnetomotive force of 1800 AT
is given to the permeability modulation winding. As can beseen in the figure, it is confirmed that the soft magnetic metal isentirely magnetized around 5000 A/m. Soft ferrite (JFE steel
MB1H) is employed as the soft magnetic metal in thisinvestigation. The initial relative permeability of MB1H is1600, and its saturation magnetic flux density is 0.5 T. Whenthe magnetic field intensity of the MB1H is 5000 A/m, its
relative permeability decreases down to 80 dues to themagnetic saturation. Therefore, it is possible to vary the

Rotor

U

modulation described above, which makes it possible to control
the leakage magnetic flux between the PM poles. The figure
shows cross sections of the 4-pole IPM rotor, where the soft
magnetic metal is inserted between the N poles and the S polesof the embedded PMs in the rotor. As already described, thepermeability modulation of the soft magnetic metal is carriedout by using the differently allocated special winding in thestator in addition to the standard three-phase windings. As
shown in Fig. 2(a), when the soft magnet metal is notmagnetically saturated, short circuits are caused between the
PM poles of the rotor. Therefore, the main magnetic flux of thePMs hardly interlinks to the stator windings, which isequivalent to the field weakening operation. On the other hand,since the reluctance between the PM poles is increased due tothe magnetic saturation of the soft magnetic metal, most part of
the PM flux interlinks to the stator windings, whichcorresponds to the intensified field operation as illustrated inFig. 2(b). As described above, the proposed strategy achieves
the adjustable magnetic flux operation by means of
permeability variation caused by the magnetic saturation of thesoft magnetic metal placed between the PM poles of the rotor.
Many of the adjustable magnetic flux approaches give themotor some kinds of electromagnetic energy to weaken the PM
flux of the rotor as presented [1]-[5], but the proposed strategy
does not energize the rotor for the field weakening operation,
which is completely different from any other approaches in thepast. The magnetic saturation of the soft magnetic metal iscontrolled by a radial direction magnetic flux, which is a DC
magnetic flux and is generated by a zero-phase current.
Therefore, in order to have a magnetic flux path for thepermeability modulation, a three-dimensional magnetic circuitdesign of the motor is required.

4000
3000
2000
1000

Rotor core0

Fig. 4. Magnetic field distribution causedby permeability modulation coil.

permeability of the soft magnetic metal placed between the PMpoles by using the magnetomotive force of the permeability
modulation winding.

3.5

6

E0rms=1.64 V

E.m.f. (V)E.m.f. (V)

2.5

2
0
-2

2.0
1.5
1.0

-4
-6

0 AT
1800 AT
Air

3.0

4

0.5
0

60

120

180

240

300

0.0

360

Electrical angle (deg)(a) Without magnetomotive force for permeability modulation.

0

1

2

3

Order

4

5

6

(a) Back e.m.f.

6

E0rms=2.33 V

0.35

4

0 AT
1800 ATAir

2Torque (Nm)

E.m.f. (V)

0.30

0
-2
-4
-6

0

60

120

180

240

300

0.25
0.20
0.15
0.10
0.05

360

0.00

Electrical angle (deg)(b) With magnetomotive force for permeability modulation.
Fig. 5. No-load electromotive force waveforms.

1

2

3

Order

4

5

6

(b) PM torque.

Fig. 6.

FFT analysis results of back e.m.f. and PM torqueof principle motor model.
0.40

Permeability modulation m.m.f. 0 AT
Permeability modulation m.m.f. 1800 AT

0.35
0.30

Torque (Nm)Figure 5 shows no-load induced electromotive force (e.m.f.)
waveforms at the rotating speed of 1800 r/min in the two casesof 0 AT and 1800 AT of the magnetomotive force of thepermeability modulation winding. Figure 6 shows the FFTanalysis results of the no-load induced e.m.f. and theelectromagnetic torque when 600-AT armature magnetomotive
force is given to the q-axis winding. According to Fig. 5 andFig. 6(a), the fundamental component of the induced e.m.f. isincreased by 30 % when the magnetomotive force of thepermeability modulation winding is 1800 AT, compared with
the 0-AT case. In addition, the average torque is also raised by
17 % as can be seen in Fig. 6(b). Hence, it is found that themagnetomotive force of the permeability modulation winding
makes it possible to adjust the fundamental component of theinduced e.m.f. However, even-number order harmonics aresuperimposed to the no-load e.m.f. as shown in Fig. 6(a).
These harmonic components are caused by the DC magnetic
flux used for the permeability modulation. The DC magnetic
flux intensifies either N poles or S poles, and weakens the restof the poles, which causes imbalance of the PMs like
consequent pole motors. Figure 7 shows current phase angle
and torque characteristics under the condition of 600-AT
magnetomotive force in the stator windings. There exists the
maximum torque per ampere (MTPA) operating point at thecurrent phase angle around 30 deg., which proves that the
proposed motor has the inverse saliency.

0

0.25
0.20
0.15
0.10
0.05
0.00
-0.05
-90

-60

-30

0

30

60

90

Current phase (deg)Fig. 7.

Current phase-torque characteristics.

Three-phase windings

1
E
2 dc

1
E
2 dc

Back e.m.f.

ia + 1⁄3 io

Zero-phase windingib + 1⁄3 io

io

ic + 1⁄3 io

Fig. 8.

Circuit configuration of proposed motor drive.

DC

DC

AC

DC

AC

AC
AC
DC

Rotor

DC
StatorDC(a) Method using DC/DC converter.

(b) Method using time and/or space harmonics.

(c) Method using zero-phase current.

Fig. 9. Motor drive circuits.
TABLE Ⅰ.

Evaluation of various motor drive circuits.

Motor drive circuits

Switching loss

Controllability

Power supply voltage utility

Method using DC/DC converter

☓

○

○

Method using time and/or space harmonics

△

☓

○

Method using zero-phase current

○

○

○

III. GENERATION OF MAGNETIC FLUX FOR PERMEABILITY
MODULARION
A. Proposed Circuit Configuration of Motor Drive
The proposed circuit configuration is depicted in Fig. 8,
where the adjustable magnetic flux PM motor is connected to athree-phase full-bridge two-level inverter with 4 wires.
Therefore, it is possible to control not only the three-phasecurrents but also the zero-phase current. Since the proposed
configuration gives three-degrees of freedom of the motor
current control, the zero-phase current can be controlled
independently of the conventional d-axis and the q-axis
currents in the vector control algorithm. The zero-phasecurrent is utilized for the permeability modulation of the soft
magnetic metal in the rotor. The voltage equation of theadjustable magnetic field PM motor is as follows: v0  ( Ra + 3R0 ) + 3pL0  0
vd  = 
 vq  0  0Ra + pLd
ω Ld

  i0   0 
  
−ω Lq  id  +  0 
Ra + pLq   iq  ωΨ f 
, (1)0where the variables and the parameters are defined asv0, vd, and vq: voltages on the 0-d-q reference frame,i0, id, and iq: currents on the 0-d-q reference frame,
Ra: three-phase winding resistance,
R0: zero-phase winding resistance,
L0, Ld, and Lq: winding inductances on the 0-d-q reference
frame,
p: differential operator, andω: angular speed.

B. Comparison among Zero-Phase Winding Excitation

Methods
Figure 9 and TABLE Ⅰ indicates 3 examples of zero-phasewinding excitation methods, which are used to generate thepermeability modulation magnetic flux. The table includes
qualitative evaluation results from the viewpoints of switchinglosses, controllability, and power source voltage utility. The
method to use a DC/DC converter introduced in (1)-(4) has
good controllability because it can control the field magnetic
flux independently of the d-axis and the q-axis currents. It also
has good performance in terms of the power source voltage
utility because the separate DC/DC converter can apply its
voltage to the zero-phase winding regardless of the main
inverter DC bus voltage.
However, additional power
semiconductor devices are definitely indispensable for the
DC/DC converter circuit, which leads to the extra power losses
such as the switching loss and the conduction loss. On theother hand, if the zero-phase winding is self-excited like (4)(9), it is impossible to control the permeability modulation
magnetic flux freely regardless of the operating points of the
motor. The method is based on use of space harmonics
generated by the concentrated windings of the motor, and it
requires diode rectifiers on the rotor to obtain the zero-phasewinding excitation. The space harmonic magnetic flux induces
the e.m.f. proportional to the rotating speed, so the zero-phasecurrent is almost proportional to the rotating speed, which
implies that this method is not applicable to the proposedadjustable magnetic flux PM motor.
The proposed circuit configuration using the three-phasefour-wire system does not require extra power semiconductor
devices at all, and makes it possible to control the zero-phasecurrent independently of the d-axis and the q-axis currents. In
addition, the potential level at the neutral point of the motor
does not vary because the zero-phase current is DC. The zerophase winding can be wound separately from the three-phasewindings, so the number of turns can be determined regardlessof the three-phase windings, which implies that the

6
x

Rotor

4

Back e.m.f.
with 0 AT (V)zStator

2
0
-2
-4
-6

Soft magnetic metal(a) Principle motor model

Magnetic steel sheet

Permeability
modulation windng
Stator
Rotor

Back e.m.f.
with 1800 AT (V)Soft magnetic composite

6
4
2
0
-2
-4
-6
0

60

120

180

240

300

360

Electrical angle (deg)(a) Principle motor model.
(b) Split rotor model

superimposed zero-phase current onto the three-phase currents
can be reduced without sacrificing the magnetomotive force.
IV. INPROVED MAGNETIC CIRCUIT DESIGN TO REDUCE
HARMONICS IN ELECTROMOTIVE FORCE.
A. Split Rotor Structure and Its Magnetic Circuit Design
The principle model of the adjustable magnetic flux IPM
motor has a distorted e.m.f. waveform as shown in Fig. 5,
which is caused by the even-order harmonic components. In
order to reduce the even-order harmonic e.m.f., the zero-phasewinding to generate the permeability modulation magnetic fluxis placed in the center of the stator and between the split rotor,
surrounding the shaft. Fig. 10(b) illustrates the split rotor
model, which has an advantage in simplification of the motor
configuration. The most important merit of this approach issignificant suppression of the low-order harmonic e.m.f.,
especially the 2nd and the 4th. For fair comparative evaluation
between the principle model and the split rotor model, only thestator shape is changed, but the rotor volume, the usedmaterials, the winding turns, the PM volume, etc. are identicalwith each other.
B. Performance Evaluation of Adjustable Magnetic Flux
Operation
Figure 11(a) shows no-load e.m.f. waveforms of theprinciple model at the rotation speed of 1800 r/min under theconditions of 0-AT and 1800-AT magnetomotive force of thezero-phase winding. Figure 11(b) also shows no-load e.m.f.
waveforms of the split rotor model at the same speed, but themagnetomotive force of the zero-phase winding is 0 AT and
900 AT. Figure 12 is FFT analysis results of the no-load e.m.f.
of the two models. As can be seen in these figures, thefundamental components of the no-load e.m.f. have been varied
by 40 % depending on the magnetomotive force of the zerophase winding. The most important different point between the

6
4

Back e.m.f.
with 0 AT (V)Cross section of principle and split rotor models.

2
0
-2
-4
-6

Back e.m.f.
with 900 AT (V)

Fig. 10.

6
4
2
0
-2
-4
-6
0

60

120

180

240

300

360

Electrical angle (deg)(b) Split rotor model.

Fig. 11.

Waveforms of back e.m.f.

two models is the low-order harmonic components. It is
confirmed that the 2nd and the 4th harmonics are effectively
suppressed. The split rotor model intensifies either of the Npoles or the S-poles in the upper rotor due to the DC zero-phasemagnetic flux, but the other poles are intensified in the lower
rotor. Therefore, the total magnetic flux gained by all the Npoles and all the S-poles is balanced, which results in theefficient suppression of the even-order harmonics in the no-lad
e.m.f. Moreover, the split rotor model can achieve almost same
performance of the magnetic flux adjustment even with 900AT zero-phase magnetomotive force. This advantage is
obtained by a common magnetic circuit of the zero-phasemagnetic flux between the two rotors.
Figure 13 shows electromagnetic torque waveforms at 1800
r/min under the condition of 600-AT q-axis magnetomotive
force, and their FFT analysis results are indicated in Fig. 14.
The average torque is almost same between the two models,
but the torque ripple is efficiently reduced by the split rotor
model.
Particularly, the 3rd harmonic component isdramatically reduced by 87 %. The effective reduction of theeven-order components of the no-load e.m.f. contributes great

3.0

3.0
0 AT
1800 AT

2.5
Back e.m.f. (V)Back e.m.f. (V)

2.5
2.0
1.5
1.0

2.0
1.5
1.0
0.5

0.5
0.0

0

1

2

3 4
Order

5

0.0

6

(a) Principle motor model.

Fig. 12.

2

3 4
Order

5

6

0.45
0 AT
900 AT

0.40

0.35

Torque (Nm)Torque (Nm)1(b) Split rotor model.

0 AT
1800 AT

0.40

0.30
0.25
0.20

0.35
0.30
0.25

The proposed strategy gives three-degrees of freedom inthe current control of the IPM motor owing to the zero-phasewinding separate from the ordinary three-phase windings. Thismeans that the magnetic flux control can be carried out by not
only the zero-phase current but also the conventional d-axis
current, and that the torque can be controlled by the zero-phasecurrent as well as the conventional q-axis current. Therefore, itis necessary to consider the extended MTPA control and theextended field weakening control, taking the permeability
modulation, i.e., zero-phase current control, into account.

0.20

REFERENCES

0.15

0.15
0

60

120

180

240

300

0.10

360

[1]

0

60

120

180

240

300

Electrical angle (deg)Electrical angle (deg)(a) Principle motor model.

(b) Split rotor model.

Fig. 13.

0.30
0 AT
1800 AT

0.20
0.15
0.10

0 AT
900 AT

0.25
Torque (Nm)

0.25

0.20
0.15
0.10
0.05

0.05
0.00

360

Comparison of torque waveforms.

0.30

Torque (Nm)0Comparison of FFT analysis results of back e.m.f.

0.45

0.10

It has been confirmed that the adjustable range of the noload e.m.f. is 40 % in both of the two motor models eventhough the both models employ low-residual-flux-density
ferrite magnets. Further improvement of the adjustable rangeof the magnetic flux is expected through the optimization of themagnetic circuit, shape and material modification of the soft
magnetic metal, use of higher-grade PM, and so forth.

0 AT
900 AT

0

1

2

3
4
Order

5

6

(a) Principle motor model.

Fig. 14.

0.00

0

1

2

3 4
Order

5

6

(b) Split rotor model.

Comparison of FFT analysis results of torque.

suppression of the 3rd harmonic component of the torque in the
split rotor model.
V. CONCLUSION
A novel strategy to achieve an adjustable magnetic flux
operation of the IPM motor has been described in the paper,
which is based on a permeability modulation technique. Detailof the magnetic circuit design and performance of theadjustable magnetic flux operation have been discussed, taking
up two models of the adjustable magnetic flux IPM motors, i.e.,a principle model and a split rotor model. By taking advantageof the permeability modulation technique, the adjustable
magnetic flux operation has been achieved by the different
approach rather than conventional approaches without
sacrificing inverse saliency based reluctance torque.
Furthermore, utilization of the zero-phase current of theinverter for the permeability modulation makes it possible toseparate the e.m.f. control from the conventional d-axis and qaxis current control. This zero-phase approach has significant
advantages from the viewpoint of semiconductor device counts,
inverter losses, circuit configuration, and more.

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