Isolated Three-phase AC to DC converterwith Matrix Converter Applying Wide
Output Voltage Operation
Jun-ichi Itoh
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
itoh@vos.nagaokaut.ac.jp

Satoshi Nakamura
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japans_nakamura@stn.nagaokaut.ac.jp

Shunsuke Takuma
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
takuma_s@stn.nagaokaut.ac.jp

Hiroki Watanabe
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japanhwatanabe@vos.nagaokaut.ac.jp
Abstract— This paper proposes an expanded method of theoutput voltage control region of an isolated AC-DC converter
utilizing multi-current mode operation for rapid battery
charger applications. The rapid battery chargers require wide
DC voltage regulation depending on the battery condition. On
the other hand, the AC-DC converter which is composed by adual active bridge matrix converter has the limitation of theoutput DC voltage range due to the current mode such as thecontinuous current mode (CCM) and discontinuous current
mode (DCM). In this paper, the relationship between the output
power and the output DC voltage at each current mode isclarified by the equations. In addition, the expansion of theoutput DC voltage region is achieved by changing current modecondition. Finally, the experimental results demonstrates thevalidity of the proposed method. As a result, both CCM andDCM operation are evaluated when the boost ratio is changed
from 0.7 to 1.2. In addition, the sinusoidal input current
waveform was obtained by the proposed control.
Keywords— three phase AC-DC converter, Matrix converter,
rapid battery charger

I. INTRODUCTION
Recently, the electric vehicles (EV) and the plug-in hybrid
vehicles (PHV) may become a trend of automotive
technologies for environmental problem solution, and therapid battery chargers are the key component to enhance theconvenience for users. The rapid battery chargers have thefollowing requirement, e.g., high conversion efficiency,
galvanic isolation, miniaturization. In particular, the reductionof the weight and the size is important for easy installation anddissemination. Typically, the rapid battery charger consists ofthe three-phase PWM converter and the bidirectional isolated
DC-DC converter such as the Dual Active Bridge (DAB)
converter, and it has good performance for the PFC capability
with low harmonic distortion [1]-[4]. However, the DC-link
capacitor between the PWM rectifier and the inverter has ashort life-time because an electrolytic capacitor is usually
selected for DC-link capacitor. In addition, the initial charging
circuit hinders miniaturization. On the other hand, thebidirectional AC-DC converter which has matrix converter
and the medium-frequency transformer has been studied [5]-XXX-X-XXXX-XXXX-X/XX/$XX.00 ©20XX IEEE[11]. These circuit directly convert the three-phase AC tomedium-frequency single-phase AC without the PWM
converter and the Voltage Source Inverter (VSI). Therefore, anumber of conversions is reduced to improve the conversion
efficiency. In particular, it has advantage for the size reduction
because a large inductor such as the grid-tied inductor andDC-link capacitor does not necessary.
Some control method for the matrix converter integrated
with the isolated AC-DC converter has been proposed [5]-[11].
In Ref. [5], the optimal switching pattern has been proposedto achieve the minimum current ripple on the DC side.
However, the input current still distorts by the incomplete
cancelation of the current ripple. In addition, the input current
ripple increases compared with the conventional control
scheme. In Ref. [6], the control method using DCM is
mentioned. The advantage of this method is that the control
algorism is easy owing to the simplification of the duty
calculation by DCM. On the other hand, the output voltage
control is necessary in order to regulate the battery voltage,
and the voltage feedback control is implemented to the typical
DAB converter in this system. Likewise, the isolated AC-DC
converter with the matrix converter also requires the output
voltage control for the battery voltage management. However,
the output voltage region depends the power transferable , andthe power transferable area depends CCM and DCM
associated with the buck-boost inductance value at the eachstep-up ratio. Therefore, most of the conventional methods areapplied to only the buck mode.
In this paper, the expansion method of the output voltage
regulation range for the isolated AC-DC converter with thematrix converter is proposed. The originality of this paper isthat the clarification of the output voltage range using therelationship between the output power and the output DC
voltage at CCM or DCM current mode. In particular, thecontrol method combined with CCM and DCM is introduced
in order to expand the output voltage range. Finally, In theexperimental result, the rated operation (1 p.u. = 6 kW) is
demonstrated that the input current THD was less than 5% at
the boost ratio from 0.7 to 1.2. The wide output voltage control

Figure 1 shows the circuit configuration of thebidirectional isolated AC-DC converter. This circuit is basedon the three-phase to single-phase matrix converter and theDAB converter, and the matrix converter is installed on the
primary side instead of the H-bridge inverter. The matrix
converter directly converts the grid frequency to mediumfrequency AC in order to minimize the transformer. In thiscircuit, a large inductor such as the grid-tied inductor and theDC output inductor does not necessary. Therefore, it ispossible to reduce the system volume in comparison withtypical circuit using BTB configuration.

vac

 1 x  NVdc
v I v I
sign ( x )  x 1 1 2 2........... (1).
x

NV

1
I1  I 2
dc
The average current Iave during point A to B is defined as theequivalent transformer current reference Iave* in the poweringmode. Each duty of I1 and I2 expressed asID1  *1  sign( x ) Dz ................................................. (2),
I ave
D2 

I2
................................................................... (3),
*
I ave

Sps

Spt

Srp

Ssp

Stp

idc
ipr

Srn

S3

vse

N1

Stn

Ssn
Sns

S1

L

vpr

Snr

Vdc

N2

S2

Snt

S4

PWM rectifier

3P -1P matrix converter

Fig. 1. Bidirectional isolated AC-DC converter with matrix converter.

vb

010
V
100 3

V2

010
001
V2

Sector II
Sector
I2 Sector I
Iin* 100
III
V1 001
I1 '
Vz
I1
v

001
V
100 4

Figure 2 shows the fundamental voltage vector diagram
and the current vector Iin*. In the matrix converter, in order to
output the input current command in a sinusoidal shape, space
vector modulation (SVM) is used so that the average output
current in each switching cycle is constant. V1 is the maximum
phase voltage, and V2 is the intermediate phase voltage of theline voltage. In addition, I1 and I2 are two vectors close to thecurrent command vector and have a cycle of six times the gridvoltage. Outputting the input current command Iin*, the input
current is controlled by outputting I1 and I2 during V1 and V2
vector output. In order to obtain the medium-frequency
voltage, V1 and V2 are output in the first half of the cycle, andthen the inverse vectors V4 and V5 are output in the second halfof the cycle.
However, In the bidirectional isolated AC-DC converterwith matrix converter, the leakage inductance of thetransformer smoothes the current. As a result, the inductancevalue is very small that a large ripple current occurs within theswitching cycle. This problem increases the distortion of the
input current and output current. Therefore, this converter isnecessary to eliminate the ripple current. Figure 3 shows amedium-frequency current waveform with the current ripple
cancellation method [7]. The ripple cancelation method isachieved by the selection of suitable output vector which is
employed by the conventional SVM. The amplitude of thecurrent vector during the power transmission period is madeequal to the medium-frequency current command value Iave*by changing the output order of the negative voltage vector.
As a result, the ripple current is equivalently neglected in theswitching cycle, therefore the distortion of the input current
and the output current without ripple can be achieved. The
operation modes are separated into two modes such as buckmode and boost mode depending on the grid voltages and DC
voltage. In addition, the output vector is selected with theconsideration of the minimization of the reactive current. The
sign function is defined as

vu iu
vv iv
vw iw

Input filter

II. CIRCUIT CONFIGRATION AND CONTROL METHOD

Spr

LC filter

is possible by changing the current mode according to theboost ratio and output power.

Sub
sector B

SectorIV

Sector
VI

I2'

010
001

Sector I

Sub
sector A

100
V1
001
Vz : Zero vectors
001
Vz : Zero vectors
V5
V6
010
V7 100 V8 010 V9 001
100
010
001
Ex.) V1 100 Srp,Spr = ON, Ssp,Sps = OFF, Stp,Spt = OFF
001 Srn,Snr = OFF, Ssn,Sns = OFF, Stn,Snt = ON
※The number of ”1” means switch is turn on.
Sector V

(a)
200

Vz

Space vector modulation
vu

Phase
voltage 0vvvw[V] 200
VA

SectorLine 300
voltage 200

VIA VIB
IA

IB

IIA

IIB

IIIA IIIBIVA IVB

v1
v2

[V]

100
Current 20

reference

VB

I1
I2

[A]

0

(b)

Relationship between grid voltage and current reference
Fig. 2. Fundamental space vector diagram.

0

0.5

1.0

1.5

2.0 [ms]

where Dz is the compensation duty for D1. The current during
Dz is a reactive or active current in the buck or boost mode.
Therefore, to compensate for the effect of the current in Dz,
an addition or subtraction from D1 is required and isexpressed as(v1  NVdc )1
T ( D  Dz (1  sign( x )))L2............... (4)(v1  NVdc )(v2  NVdc )
*

TD1 
TD2  2 I ave
L
L
where L is the leakage inductance referred to the primary sideof the transformer, Vdc is the DC voltage. In the equation (3),
Da is expressed as2D 

2 I ave*L
 (v1  NVdc ) D1  (v2  NVdc ) D2
T
2(v1  NVdc )1
Dz (1  sign( x ))2

(5).

Dz is decided to equalize the amount of the change during T1
to T2 respectively.

Dz  sign ( x )(v1  NVdc ) I1  (v2  NVdc ) I 2... (6)*
2 NVdc I ave

Substring (6) into (1), D1 is expressed by I (V2  Vout ) D2  (V1  Vout )D1   *1............. (7) I ave V1  Vout  2Vout
Finally, the zero current period D0 is defined byD0  1  (2D  D1  D2  Dz ) ........................ (8)ipr 0

The equivalent transformer current reference Iave* is restrictedby the switching frequency, the leakage inductance L, the gridvoltage, and the DC voltage. The equivalent transformer
current reference is expressed as

I ave* 

v1
v2
vpr

B’

T1

T2 TzTa(a)

Buck boost state

T0 Ta T2

T1

C’

TzTa T0

B

Iave
ipr

3
T
Pb _ buck ( Vac 2  Vout 2 ) ....................... (10),L
2 6Vac 2

0

A

C
O

O
C’

T0 Ta Tz

T1

T2 Ta

T0

B’

Ta Tz T2

A’
T1

Ta T0

(b) Boost stateFig. 3. Ripple cancellation method in order to simplify the duty
calculation and minimize the reactive power.

 vac

N 2Vdc 2  2vac 2....................... (11)L
2 2 NVdc

where Vac is the RMS value of the grid voltage. The equations(10) and (11) express the boundary condition without CCM
during 0 to 30 degree completely. Solving (9), the maximum
output power Pout is expressed as

 6 vac NVdc....................................... (12)8LChanging (12), the leakage inductance is calculated byusing (13).
(13)

Figure 4 shows the relationship between the output power
and the ratio of the input and output voltage. CCM and DCM
are used to realize the boost and buck operation. According
to (12), the output power is restricted by the switchingfrequency, the leakage inductance, the grid voltage, the DC
voltage, and the turn ratio. The DAB converter for the DCDC conversion has the similar limitation by the same circuit
parameter. The medium-frequency current has zero current
period in DCM when the DC voltage is significantly larger orlower than the grid voltage at a light load.

Boost ratio is extended
using both CCM and DCM

1.4

394.8

Boost ratio NVdc / 2Vac

O

0

-v2
-v1
Vdc
vse 0
-Vdc

 6 vac NVdc....................................
16
P

C

O

Active mode Circulating mode Active modev1  NVdc T1
 K1  K 2  . (9)(1  D0 )
2
2
L


Vout

L

A

T 0 TaThe current references for CCM and DCM are sameat the boundary condition. The output power at theboundary condition is expressed as

P

B

A’

 v  NVdc T 
K1   12Lv  NVdc I1 (v1  NVdc )  I 2 (v2  NVdc ) T
K2  1
2
NVdc
L

Pb _ boost Circulating mode
Active mode
Active mode

v1
v2
vpr 0
-v2
-v1
NVdc
vse 0
-NVdc
i +i i2
Iave= 0 2 i
2 0Boost ratio becomes 1.0 DCM
when NVdc is 282 V

1.2

338.4

282
1.0

CCM
Boost ratio is 0.6 to 1.0 by
Eq. (11) CCM when p is 0.6 p.u.
out

CCM+DCM

0.8

225.6

Eq. (12)

0.6

169.2

DCM

0.4

112.8

Eq. (10)56.4
0.2

Pout: 1 p.u. = 6kW
Boost ratio: NVdc / 2Vac
L= 21 mH(%Z=0.01%)
fsw= 50 kHz

No operation region

0
0

0.2
1.2

0.4
2.4

0.6
3.6

0.8
4.8

1.0
6

Output power Pout[p.u.]

Fig. 4. Relationship between output power and boost ratio. The operation mode and current mode are changed by the grid voltage, DC voltage and transition
power. Each line means the boundary condition to operate the proposed converter at CCM and DCM.

III. COMMUTATION ERROR COMPENSATION METHOD ON
RIPPLE CANCELLATION METHOD
A. Voltage error caused due to 4-step voltage commutation
The matrix converters need switching that prevents shortcircuits in the power supply and does not open the load withinductive loads. As a method of commutating, whilesatisfying this condition, there is a voltage commutation
method that commutates depending on the magnitude of thevoltage and a current commutation method that requires the
direction of the load current. However, the load current has amedium-frequency, the current commutation method requires
a highly accurate current sensor, which increases the cost.
Therefore, in this study, we apply a 4-step voltage
commutation that does not require a current sensor.
Figure 5 shows the model of voltage commutation using abidirectional switch and the switching sequence. When thecurrent direction is positive (iload > 0) and commutation ismade from the power supply va to vb, consider the case where
S1a and S1b are ON and S2a and S2b are OFF under theconditions of va > vb. At this time, the switching sequence is
executed in the order of S2a:ON, S1a:OFF, S2b:ON, S1b:OFF.
At this time, commutation from va to vb occurs when S1a inthe second step is turned off. That is, the output voltage isdelayed from the ideal timing by the dead time Td. Similarly,
when commutating from the power supply va to vb under thecondition of va < vb, the commutation from va to vb occurs in
the third step, so the actual output voltage is delayed by 2 Td.
Therefore, the delay of the output voltage due to commutation
is not always constant and is determined by the voltage
condition and the current direction, so it is necessary tocompensate for the duty considering the delay.
B. Voltage commutation compensation during DCM
operation under boost condition
Figure 6 shows the operation waveforms of ideal
commutation and voltage commutation in DCM. the voltage
magnitude relationship is vu > vv > vw. In the ripple cancel
method, the drive mode as DCM in the light load region and
CCM in the heavy load region. During the powertransmission period, duty is generated so that the amplitudeof the current vector becomes equal to the medium-frequency
current command value iave* in each cycle. However, voltage
commutation causes a delay in the output voltage due tocommutation. Fig. 6(b) shows the outline of operationconsidering the delay of the output voltage due tocommutation. For example, when commutating from T0 (V9)to Tb (V4), the voltage is commutated from the w phase to theu phase, which is the commutation from the minimum voltage
phase to the maximum phase. Since the current direction ispositive, the commutation delay is 2Td. Then, whencommutating from Tb (V4) to T2 (V5), it is commutation of thevoltage from u phase to v phase, and it is commutation fromthe maximum voltage phase to the intermediate phase. Since
the current direction is positive, the delay due to thecommutation of the voltage is Td. In other words, the output
voltage period of Tb(V4) is shortened by Td due tocommutation. As a result, the current average value iave does
not match the command value. In particular, since theinductor voltage during the Tb period is higher than in otherperiods, the error due to the dead time greatly affects the error
between the transmitted power and the command. Moreover,since the voltage during the ideal period is not output, themedium-frequency initial current is not held at zero. This
causes distortion in the input current because it does not meet

1

S1a Negative

va
vb

S1b
S2b

2

3

4

S1a
S1b
va > vb S2a
S2b

va
vb

or
Positive

Td

error

S1a
S1b
va < vb S2a
S2b

iload

S2a

va
vb

error

Td

Fig. 5. Commutation model and voltage error.
Active mode Circulating mode Active mode
Tb T2
T2
T1
T0 Tb T1
T0
T0
(V9) (V1) (V1) (V2) (V9) (V4) (V5) (V4) (V9)MC output voltage vpr [V]0
PWM rectifier
output voltage vse [V] 0

iave*
0Transfer primary current ipr [A](a) Ideal commutation
Active mode Circulating mode Active mode
T2
Tb T2
T1
T0 Tb T1
T0
T0
(V9) (V1) (V1) (V2) (V9) (V4) (V5) (V4) (V9)0MC output voltage vpr [V]

Td

Td

Td

0
PWM rectifier
output voltage vse [V]Offset occurs

iave(iave < iave*)0
Transfer primary current ipr [A]Commutation delay

2Td 0

Td 2Td2Td Td2Td

2Td(b) Voltage commutation
Fig. 6. Comparing the operation modearound the DCM operation.

the DCM duty calculation assumptions. Therefore, we
consider a method of compensating for voltage errors due tocommutation.
Regarding the voltage error compensation amount during
DCM operation, the error in the output voltage due tocommutation is shortened by Td during the period of Tb andextended by Td during the period of T2as shown in Fig. 7(b).
When expressed by, it becomes equation (14).
'Tb  Tb  Td...................................................................(14) 'T2  T2  Td

C. Voltage commutation compensation during CCM
operation under boost condition
In the ripple cancellation method, the drive mode as DCM
in the light load region and CCM in the heavy load region.
However, the system changes at 300 Hz, so the duty alsochanges at 300 Hz cycles. Depending on the operating
conditions, a period in which DCM and CCM coexist at a 300
Hz cycle occurs.

Ta
Tb



I ave* 1
K cof 1  ( K cof 12  K cof 2 )2

K cof v1  NVdc T
2L ........................ (15)Ta Tb
vpr [V]0vse [V]0vse [V]0ipr [A]0

ipr [A]
0

Further, the calculation formula of each duty is expressed by
the formulas (16) to (21) using the current command iave* ofthe formula (15).
(V  Vout ) D2  I
 (V1  Vout )T1   *1  2.......................... (16)

I
V

V


ave
1
out

 2Vout
T2 

I2..................................................................... (17)I ave*( NVdc  v1 ) I1
1
Tz  (
 T2 ( NVdc  v2 ))I ave*
2 NVdcT ................. (18)*
2 I ave
Lf sw  (v1  NVdc )T1  (v2  NVdc )T2
 Tz ........ (19)2(v1  NVdc )T1

T2 Ta Tb

T2 T1

Ta

Current direction inversion occurs(a) Around the DCM(b) Heavy load area
Fig. 7. Outline of the CCM operation.

Table 1. Experimental parameters.
Quantity
Rated power
Three-phase AC voltage
DC voltage
Input frequency
Carrier frequency
Leakage inductanceTurn ratio of transformer

SymbolP
vac

Dead-time
vpr[500V/div]

4[msec/div]

0

Value
6.0 kW
200 V
Max 100 V
50 Hz
50 kHz21 µH
3.3:1
0.19 mH
10 mF
250 ns

Vdc
f
fsw
L
N1:N2
Lf
Cf
Td

Input filter

vpr[500V/div]

4[msec/div]0
vse[100V/div]0

vse[100V/div]0
ipr[25A/div]0

ipr[25A/div]0
Offset is occuredv  NVdc  I1 (3 NVdc  v1 )  I 2 (3 NVdc  v2 )T
K cof 2  1
2
NVdc LT1

Ta

vpr [V]
0

Figure 7 shows the CCM operation waveforms in theregion close to DCM and the heavy load region. In Fig. 7(a),the output period of Ta is shorter than the dead time, and the
problem occurs that the medium-frequency current direction
reverses before the voltage vector switches. When the deadtime error compensation is applied, the current direction inwhich the voltage vector switches before and after
compensation is different, and the voltage error due tocommutation changes the compensation amount before andafter compensation. Further, when the inductor current
becomes zero during the commutation period, a voltage
polarity reversal phenomenon occurs in which the polarity ofthe transformer voltage is reversed. From the above, it is notpossible to compensate for the output voltage error due to theregion in which the current direction is reversed [12].
Therefore, the proposed method of compensating for the
nonlinear power error due to the voltage polarity reversal
phenomenon by using the voltage waveform including thezero voltage period. The current direction reversal
phenomenon is suppressed by providing a zero current period
in the inductor current and controlling the zero current
periods beyond the dead time. The current command iave* isexpressed by equation (15).

Ta
T 2 Tb T 2

T1

(a)

Offset is reduced

Without compensation (b) With compensation
Fig. 8. Waveforms of matrix converter and rectifier.

error compensation is possible. However, there is a drawbackthat the number of times of switching increases because thenumber of times of switching the voltage vector increases.
The voltage error compensation amount is given by equation(22) and (23). Expression (23) is the compensation amountwhen the direction of the medium-frequency current ipr isnegative.
'Tb  Tb  Td........................................................... (22) 'T0  T0  Td'
T1  T1  Td
(ipr < 0)............................................... (23) 'T2  T2  Td

Tb  Ta  Tz ................................................................... (20)

IV. EXPERIMENTAL VERIFICATION

T0  1  T1  T2  Ta  Tb  ............................................. (21)Fig. 8(a) shows the primary side voltage, current, andsecondary side voltage in the conventional DCM region, andFig. 8(b) shows the primary side voltage, current, andsecondary side voltage with power error compensation. The
transmission power command at this time is 2000 W. From
Fig. 8(a), in the conventional method, an error occurs in the
output voltage due to the commutation operation of theBy driving with DCM, the current direction when switchingeach voltage vector does not change from that of DCM usedwith a light load. Therefore, the voltage error due tocommutation is constant in the switching cycle, and powermatrix converter, and an offset occurs in the mediumfrequency current during the period when the current isessentially zero. On the other hand, as shown in Fig. 8(b), themedium-frequency current also converges to zero insynchronization with the zero voltage period by suppressing
the output voltage error by error compensation.
Figure 9 shows the input/output characteristics in the
DCM region before and after voltage error compensation. In
Fig. 9(a), it was seen that the initial current, which is thepremise of the duty calculation in the DCM, which does notsatisfy the condition of zero, and that the input current isgreatly distorted. In Fig. 9(b), it was confirmed that bysuppressing the offset that was superimposed on the mediumfrequency current, the assumption of the ripple cancellation
method that the initial current was zero was satisfied, and the
input current distortion could be reduced.
Figure 10 shows a comparison diagram of the proposed
DCM operation and the CCM operation waveform with thecurrent direction reversed. In CCM operation, as shown in
Fig. 7(a), the current direction is reversed before the voltage
vector is switched, so the power compensation becomes nonlinear and difficult to compensate. On the other hand, in the
DCM operation proposed in <3.C>, the medium-frequency
current converges to zero in the zero voltage period, so thecurrent direction does not change when the voltage vector
changes.
Figure 11,12 shows the input/output characteristics andwaveforms of matrix converter and rectifier at the rated
conditions under the boost ratio of 0.7, 0.866.1.0, and 1.2. It
was confirmed that the system voltage and current are inphase and controlled at THD 5.0% or less under each boost
condition while controlling at a power factor of 1. Also, it isclear that the theoretical medium-frequency voltage is
generated and the slope of the current is controlled.
A ripple component of 300 Hz cycle generates on the
output current near the zero current of each phase current.
The duty of T2 that outputs the intermediate phase always
zero at the timing when the current of each phase becomes
zero, and there is a range in which the pulse width is shorter
than the dead time. As a result, a pulse drop is generated near
the zero current of each phase, and as a result, the voltage
vector that should be output is not output. This is because theduty of T2 that outputs the intermediate phase is always zeroat the timing when the current of each phase is zero, and there
is a range in which the pulse width is shorter than the deadtime. As a result, a pulse drop generates near the zero currentof each phase, and the voltage vector to be output is not output.
Figure 13 shows the THD characteristics. The THD at the
rated value is below 5% under any boost condition. However,
the input current distortion still tends to be large at light loads.
This is because the range of pulse deficiency in the period T2during which the intermediate phase is output becomes wider
at light load, and the effect of dead time becomes larger
relative to the fundamental wave.
Figure 14 shows the efficiency characteristics. The
maximum efficiency point reaches 96.7% at 1.5 kW output
under the condition of step-up ratio 0.866. The efficiency
tends to decrease after 3 kW under any boosting condition.
This problem is solved by changing the maximum efficiency
point by optimal design of the transformer and mediumfrequency inductor.

00vu[200V/div]
iu[10A/div]

Vdc[50V/div]0
idc[10A/div]

0

4[msec/div](a)0vu[200V/div]0iu[10A/div]

Without compensation

Vdc[50V/div]0

idc[10A/div]
4[msec/div]0(b) With compensation
Fig. 9. Input and output waveforms on DCM.
vpr[250V/div]

10[msec/div]0

vpr[250V/div]

10[msec/div]0
vse[100V/div]0

vse[100V/div]0iL[25A/div]0iL[50A/div]0
Polarity inversion occurs(a)

CCM operation(b)Proposed DCM
Fig. 10. Comparing the operation modearound the DCM operation.

V. CONCLUSIONSIn this paper, the output voltage region of the isolated ACDC converter with the matrix converter is proposed using themathematical equation. The proposed method shows tooperate the power transmission with a wide output voltage
range. The originality point of the paper is to clarify theoperating output voltage range using the relationship between
the output power and the output DC voltage at each current
mode. As the result, it was mentioned that the output voltage
region under CCM and DCM is difference. In addition,transmission power error compensation to improve input
current distortion under boost conditions. By analyzing thevoltage error caused by commutation, the average value ofthe medium-frequency current during the transmission power
period is approximated to the ideal value, and the offset is
reduced in the medium-frequency current during the zerovoltage period. The experimental results, the current THD at
rated power operates at less than 5% in both step-down andstep-up conditions, and the maximum efficiency value is
96.7%.

0vu[250V/div]0

vu[250V/div]0iu[20A/div]0iu[20A/div]

Vdc[50V/div]

Vdc[50V/div]00
idc[50A/div]

idc[50A/div]

4[msec/div]

0

4[msec/div]0(a) Boost ratio 0.7.

(b) Boost ratio 0.8660vu[250V/div]0

vu[250V/div]0iu[20A/div]0iu[20A/div]

Vdc[50V/div]

Vdc[50V/div]00
idc[50A/div]

4[msec/div]

0

4[msec/div]

idc[50A/div]0(c) Boost ratio 1.0.
(d) Boost ratio 1.2.
Fig. 11. Input and output waveforms of matrix converter with rated operation
10[msec/div]

vpr[250V/div]00
vse[100V/div]0

Boost ratio 0.7.
(a)

0

100

Efficiency [%]

95
95

55

0.2
1200

0.4
0.6
2400
3600
Output power Pout [p.u.]

Fig. 13. THD characteristics.

0.8
4800

(d) Boost ratio 1.2.

100

0.7
0.866
1.0
1.2

10
10

0

iL[50A/div](b) Boost ratio 0.866(c) Boost ratio 1.0.
Fig. 12. Waveforms of matrix converter and rectifier with rated operation.

15
15

THD [%]

0

0

Boost ratio

00

vse[100V/div]iL[25A/div]iL[50A/div]

20
20

0

0

iL[50A/div]

10[msec/div]

vpr[250V/div]

vse[100V/div]0010[msec/div]0
vse[100V/div]0(a)vpr[250V/div]

10[msec/div]

vpr[250V/div]

1.0
6000

90
90

Boost ratio
85
85

80
80

0

1200
0.2

0.7
0.866
1.0
1.2

2400
3600
0.4
0.6
Output power Pout [p.u.]

Fig. 14. Efficiency characteristics

4800
0.8

6000
1.0

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