Reduction in Radiation Noise Level for
Inductive Power Transfer Systems using Spread
Spectrum Techniques
Kent Inoue, Student Member, IEEE, Keisuke Kusaka, Member, IEEE, and Jun-ichi Itoh, Senior Member, IEEE

Abstract-- Two proposed methods of reducing radiation noisecaused by inductive power transfer (IPT) systems using spread
spectrum techniques were experimentally demonstrated. In IPT
systems for electric vehicles (EVs) or plug-in hybrid EVs(PHEVs), noise reduction techniques are necessary because theradiation noise from the IPT system for EVs or PHEVs must notexceed the limits defined by noise standards, such as the wellknown regulation set by the International Special Committee onRadio Interference (CISPR). In the proposed methods, theradiation noise from the transmission coils of the IPT system isspread in the frequency domain by randomly changing theoutput frequency of the inverter. The output frequency is selectedaccording to the generation of pseudorandom numbers. In thefirst proposed method, spread spectrum with a uniform
distribution (SSUD), the output frequency is selected from thefrequency range of 80 to 90 kHz with an even distribution. In thesecond method, spread spectrum with a biased distribution(SSBD), the output current of the inverter is considered. The
possibility of selecting an output frequency in the given range isbiased in proportion to the combined impedance from thetransmission coil and the resonance capacitors on the primaryside. In experiments with a 3-kW prototype, when SSUD and
SSBD were applied, the fundamental components of the radiationnoise were suppressed by 4.45 and 8.27 dBA, respectively, incomparison with the noise obtained using the conventional
system, which operates the inverter at a fixed frequency.
Index Terms—Inductive power transfer, Wireless powertransfer, Spread spectrum, Radiation noise, Electromagnetic
interference.

I. INTRODUCTIONI

n recent years, inductive power transfer (IPT) systems havebeen actively studied and developed [1–7]. In particular, thepractical realization of IPT systems for electric vehicles (EVs)or plug-in hybrid EVs (PHEVs) is highly desired because IPT
systems are capable of improving the usability of EVs and
PHEVs [8–12]. IPT systems transmit power using the weakmagnetic coupling between a primary coil and a secondary
coil through a large air-gap [13]. The fundamental principle
IPT systems are the same as that of a transformer. However,coupling coefficient between the primary and secondary coils
is typically in the range of k = 0.1 to 0.3, in IPT systems. In
Manuscript received December 20, 2016; revised April 4, 2017; accepted
May 18, 2017.
Kent Inoue, Keisuke Kusaka and Jun-ichi Itoh are with the Dept.
Electrical Electronics and Information, Nagaoka University of Technology,
Niigata
940-2188,Japan(e-mail:k_inoue@stn.nagaokaut.ac.jp;
kusaka@vos.nagaokaut.ac.jp; itoh@vos.nagaokaut.ac.jp).

modern IPT systems, compensation circuits, which containcapacitors and inductors, are used to compensate for the power
factor from the perspective of the power supply [1] [8] [10].
Compensation circuits reduce the copper loss that occurs in
the windings of coils and the power loss in power supply.
To use IPT systems in a practical applications, radiatedelectromagnetic noise must be suppressed [14] to satisfy
exiting standards, such as the well-known standard based on
CISPR11, a standard by the International Special Committeeon Radio Interference, because the IPT system must not affectother wireless communication systems, or electronicequipment.
A common noise reduction method is the use of a filter
circuit. In this method, a low-pass filter is connected to the
input stage of the primary coil. The filter circuit suppresses the
harmonic components of the current that flow into thetransmission coil. However, the power loss of the filter is large
in this method because the cut-off frequency of the filter
circuit must be close to the fundamental frequency. As
previous studies, suppression methods using a magnetic or
metal shield have been proposed [15–18]. The transmission
coils are surrounded by plates composed of a magnetic
material or metal. The radiation noise can be suppressed
because these shields alter the magnetic flux to produce eddy
currents. However, the eddy currents increase the power lossof the IPT system. Additionally, the presence of an aperture
for the magnetic path must be ensured in the IPT system. Thus,
the shielding can only provide partial suppression of theradiation noise. Previous studies [19] [20] have proposed anoise reduction method involving the formation of the current
that flows in the coils by using a primary converter. However,additional switches are required to reduce the radiation noise.
Previous study [21] has proposed a reduction method thatinvolves the use of a set of two inverters and two coils in closealignment. In this setup, the radiation noise is cancelled bypassing opposite-phase currents through the two coils.
However, the inclusion of two inverters and two coils
increases the cost of the IPT system.
Alternatively, spread spectrum techniques are widely usedin motor drive systems with a pulse width modulation (PWM)
inverter [22–24]. This technique is called random carrier
PWM. The carrier frequency of the PWM inverter is
continuously changed during operation to reduce acoustic
noise. By changing the carrier frequency, the frequency

components of the acoustic noise caused by winding
vibrations can be spread over wide range in the frequency
domain [22–24]. Spread spectrum techniques are also used inconverters to reduce conducted electromagnetic interference(EMI) [25–27]. In both applications, the carrier frequency ischanged to suppress acoustic or electromagnetic noise.
However, spread spectrum techniques have not been applied
to square wave inverters, meaning that they have not been
applied to IPT systems.
In this paper, two methods of reducing the radiation noisefor the IPT system based on the spread spectrum technique areproposed and demonstrated with the experimental results. In
the first proposed method, spread spectrum with a uniform
distribution (SSUD), the radiation noise is suppressed by
selecting the output frequency from a uniform probability
distribution. In the second proposed method, spread spectrumwith a biased distribution (SSBD), the output frequency is
selected from a biased probability distribution. The probability
distribution is proportional to the impedance combined fromcompensation capacitor and the transmission coil on the
primary side.
This paper is organized as follows. In section II, theelectromagnetic noise regulations for the IPT system aredescribed. The two proposed noise reduction methods are
explained in section III. Section IV describes theimplementation of the proposed methods in a 3-kW prototype.
The radiation noise of the IPT system using the proposed
methods is then evaluated. Finally, the effects of the spread
spectrum technique on the system efficiency are assessed.
II.

ALLOWABLE RADIATION NOISE LEVEL

Radiation noise [dBA/m] (at 10 m)

Fig. 1 shows the radiation noise regulations for IPT systems
with an output power of 7-kW or less [28]. The regulations are
currently under discussion for standardization. The use of afrequency range from 79 to 90 kHz is considered for IPT
systems of EVs. This regulation basically corresponds to
CISPR11 Group 2, Class B [29]. However, the radiation noisefrom the frequency range of 79 to 90 kHz must be mitigated toa maximum of 68.4 dBA/m. Moreover, the maximum
radiation noise in higher frequency bands (150–490 kHz) will
be mitigated by 10 dBA from CISPR Class B.
Furthermore, in the IPT system, a limit to the allowableradiation noise in the range 9 to 150 kHz must be imposed.
The radiation noise in all frequency bands except that from 79
to 90 kHz must be lower than 23.1 dBA/m. Similarly, theallowable limit in the frequency band from 526.5 to 1606.5
kHz is −2.0 dBA/m because this frequency band has beenused for amplitude modulation (AM) broadcasting.
It should be note that CISPR11 suggest measuring the noiseusing a quasi-peak measurement method. As stated in theregulations, not only the fundamental frequency component of
the radiation noise but also its harmonic components should be
suppressed.
III. NOISE REDUCTION METHODS BY SPREAD SPECTRUM
A. Compensation Circuits.
Fig. 2 shows the typical circuit configuration of the IPT
system with a series–series (S/S) compensation [30]. In the
IPT system for EVs, the primary coils are buried in roads orparking lots, and secondary coils are installed the bottoms ofthe cars. For this reason, the magnetic coupling between the
primary and secondary coils are weak. A large leakage
inductance attributable to this weak magnetic coupling causes
the reactive power to increase. To solve these problems,compensation circuits, such as series–series (S/S)
compensation, or series–parallel (S/P) compensation circuits,
are widely used to compensate the leakage inductance [31].
When an input voltage V1 is applied to the primary side, theprimary and secondary currents are calculated as
1 
r2  Req  j  L2 
C2 

I1 
V1 



1 
1 
2 2
r2  Req  j  L2 
   Lm
r1  j  L1 

C1 
C2 



I2 

jLm



1 
1 
r2  Req  j  L2 
   2 L2m
r1  j  L1 
 



C

C

1  
2 Fundamental

60

40
2nd 3rd
4th
5th

23.1
20

0

CISPR Class B Group2

-10
10

100

CISPR Class B
Group2 + 10 dB

1000



where Req is the equivalent load considering the rectifier; r1
and r2 are the equivalent series resistances of the primary andsecondary windings, respectively; L1 and L2 are the primary

80
68.4

V1 



Fundamental

79 kHz – 90 kHz

2nd

158 kHz – 180 kHz

3rd

237 kHz – 270 kHz

4th

316 kHz – 360 kHz

5th

395 kHz – 450 kHz

10000

Frequency[kHz]Fig. 1 Maximum allowable radiation noise for IPT systems of 7-kW or less for use in EVs (under discussion).

2

8 V2, DC 
 2 P2

Req 



where V2,DC is the secondary direct current (DC) voltage andP2 is the output power.
The compensation capacitors are generally selected tocancel out the reactive power at the input frequency. Thus, thecompensation capacitors can be calculated
C1 

1

L1 2



C2 

1

L2 2



The primary current with the compensation is given byI1 

r2  Reqr1 r2  Req    2 L2m

V1 



Because of presence of the compensation circuit, the input
impedance from the perspective of the output of the power
supply is relatively low. Thus, the input current contains alarge fundamental component.
The radiation noise is mainly caused by the current flowing
through the primary and secondary coils. For other converters,
shielding with a magnetic material or metal is an effective
method of suppressing radiation noise. However, in an IPT
system, the presence of an aperture for the magnetic path must
be ensured to supply the power. Thus, shielding provides alimited solution for the suppression of radiation noise.

The output frequency is selected at random in range of 80 to
90 kHz. In SSUD, the output frequency of the voltage source
inverter is selected from a discrete uniform probability
distribution. In contrast, SSBD involves selecting the output
frequency of the voltage source inverter from a biased
probability distribution. The probability distribution is biased
to be proportional to the combined impedance of thetransmission coil and the compensation capacitor on the
primary side. Because of this biased probability distribution,
the harmonic components of the current output from theinverter are spread.
Fig. 3 shows the probability distributions of the output
frequency of the inverter. Fig. 3 (a) is the probability
distribution of SSUD [32]. The probability distribution is adiscrete uniform distribution from 80 to 90 kHz. This meansthat each output frequency has the same probability of being
selected. The frequency parameter is discrete because thecarrier for the modulation of the inverter is generated in a freeprogrammable gate array (FPGA). The output frequency isrenewed at every carrier. Selecting the output frequency of thevoltage source inverter from a uniform distribution causes the
harmonic components of the voltage to be evenly spread. Fig.
3 (b) shows the probability distribution of SSBD [33]. The
input impedance of the IPT system depends on the frequency.
Thus, the probability distribution is proportional to theimpedance from the transmission coil and combined with thatfrom the compensation capacitor. Fig. 4 shows the harmoniccomponents of the primary current around the fundamental
frequency range (80–90 kHz). In the SSUD, the current in thecoils is not taken into the consideration because the output
frequency of the voltage source inverter is selected from theuniform distribution. Thus, the current in the coils has afrequency characteristic depending on the impedance of the
IPT system. On the other hand, considering the impedance ofr1+

B. Proposed Noise Reduction Methods.
In the methods proposed in this paper, the radiation noise isspread in the frequency domain by changing the output
frequency of the voltage source inverter in two different ways.
40

Probability distribution

35

15
0.15

30

650 Hz

25

10
0.10

20
15

5
0.05

10
5

00
79

80
80

81

82
82

83

84
84

85

86
86

87

Frequency[kHz]

88
88

89

90
90

0
91

k

L1

C2

r2
R

I2

L2

V2DC

Fig. 3 Typical system configuration of IPT systems for EVs.
20
0.20

40
40
35

Probability distribution

20
0.20

I1

C1

15
0.15

30
30
25

10
0.10

20
20

5
0.05

10
10

15

5
00
79

80
80

81

82
82

83

84
84

85

86
86

87

88
88

89

90
90

Frequency[kHz]Fig. 2 Probability distributions for the proposed spread spectrum techniques. (a) Proposed method I: SSUD. (b) Proposed method II: SSBD.

00
91

Impedance[W]and secondary inductances, respectively; C1 and C2 are theprimary and secondary compensation capacitors, respectively;
Lm is the mutual inductance; and  is the angular frequency ofthe power supply. Note that the voltage V1 is the fundamental
component of the output voltage of the inverter. The
equivalent load is given by [31]the IPT system, the SSBD selects the output frequency fromthe biased distribution. It helps further spreading the current
harmonics around the fundamental frequency in the frequency
domain in comparison with constant frequency operation and
the SSUD. Thus selecting the output frequency of the voltage
source inverter from the biased distribution is effective to
reduce the noise. Note that the probability distribution, which
is proposed in the paper, is not actually an optimum
distribution because the probability distribution was
introduced through a trial and error process.
Table I shows the assignment of the pseudorandom
numbers for the output frequency using SSUD (Table I (a))
and SSBD (Table I (b)). The output frequency is selectedaccording to the generation of 7-bit pseudorandom numbers.
Fig. 5 shows the pseudorandom number generation method.
The pseudorandom numbers are generated using a maximal
length sequence (M-sequence) [25] [34] in the digital signal
processor (DSP). Different pseudorandom number generation
methods can be used. However the generation method using
an M-sequence was selected for use in this study because acomplex generation method for pseudorandom numbers is notsuitable as an algorithm for DSP-based implementation. An
M-sequence random number is generated as the exclusive ORof XZ-p and XZ-q as shown
X Z  X Z  p  X Z  q where XZ-p and XZ-q are the present values of XZ delayed byperiods of p and q, respectively, with p > q. In this study, p = 7
and q = 1 were used. Moreover, the pseudorandom number is
a 7-bit number.
IV. EXPERIMENTAL RESULTS
A. Experimental Setup.
Fig. 6 and Table II show the configuration for the prototype
and the specifications, respectively. In these experiments, a420-V DC voltage power supply was used. SiC MOSFETs
and SiC diodes were used at switching devices. The SiC
MOSFETs are controlled by the FPGA and the DSP.
The inductances of the primary and secondary coils were
designed according to the following equation [11] assuming

the effect of the spread spectrum can be ignored;
L1 

L2 

R eq V1,2DC

 0 k 0 V 22,DCReq

0 k 0









where 0 is the center frequency of the frequency range used
for the spread spectrum. The compensation circuit can becalculated from the center frequency 0 using (4) and (5). Thismeans that the compensation circuits are designed to resonate
at 85.1 kHz.
Fig. 7 shows the primary and secondary coils for the
prototype. Solenoid-type coils [35] are used as thetransmission coils. The transmission distance is 150 mm,
assuming transmission from the road to the bottom of the EV
or PHEV.
B. Radiation Noise Measurement Conditions.
Fig. 8 shows the setup of the probe (ELECTRO-METRICS
EM-6993) used to measure the radiation noise. The radiationnoise was measured at two points, A and B, and the distancefrom the edge of the transmission coils and to eachmeasurement position was 500 mm. Both measurement points
are in the center of the transmission coils along the z-axis. At
position A, the flux, crossing the x-y and y-z planes were
measured. At position B, the flux crossing the y-z plane wasmeasured.
The measurements were conducted in a shielded room.
These measurements cannot satisfy the requirements for themeasurement of radiated emissions established by CISPR interms of the measurement environment. The standards requiremeasurements to be conducted with a 10-m test range
anechoic chamber, which eliminates unwanted noise at afrequency of 85 kHz with good performance. Thus, in thispaper, only the results of the radiation noise reduction
obtained using a conventional system and the currently
proposed methods are compared.

[A]

6.0

4.0

2.0

0
80.0

82.5

85.0

87.5

[kHz]

80.0

82.5

85.0

87.5

[kHz]

80.0

82.5

85.0

87.5

[kHz]

Fig. 4 Probability distributions for the proposed spread spectrum techniques. (a) Constant frequency. (b) Proposed method I: SSUD. (c) Proposed
method II: SSBD.

TABLE I ASSIGNMENT OF THE OUTPUT FREQUENCY. (A) PROPOSED METHOD I: SSUD. (B) PROPOSED METHOD II: SSBD.
pseudorandom
numberFrequency [kHz]pseudorandom
numberFrequency [kHz]10000001

80.00

65

1000001

85.47

⦙

⦙

80.00

⦙

⦙

8

0001000

80.00

72

9

0001001

80.65

⦙

⦙

16

pseudorandom
numberFrequency [kHz]pseudorandom
numberFrequency [kHz]1000000180.00

420101010

85.47

85.47

⦙

⦙

80.00

⦙

⦙

85.47

100100085.47

11000101180.00

47010111185.47

73100100186.21

12000110080.65

480110000

86.21

80.65

⦙

⦙

86.21

⦙

⦙

80.65

⦙

⦙

86.21

0010000

80.65

801010000

86.21

20001010080.65

55011011186.21

17001000181.30

81101000186.96

21001010181.30

56

0111000

86.96

⦙

⦙

81.30

⦙

⦙

86.96

⦙

⦙

81.30

⦙

⦙

86.96

24
25

0011000

81.30

881011000

86.96

27001101181.30

65

1000001

86.96

0011001

81.97

89101100187.72

28

0011100

81.97

66

1000010

87.72

⦙

⦙

81.97

⦙

⦙

87.72

⦙

⦙

81.97

⦙

⦙

87.72

32

0100000

81.97

96110000087.72

32

0100000

81.97

77100110187.72

33

010000182.64

97110000188.50

33

0100001

82.64

781001110

88.50

⦙

⦙

82.64

⦙

⦙

88.50

⦙

⦙

82.64

⦙

⦙

88.50

40

0101000

82.64

104

1101000

88.50

35

010001182.64

91101101188.50

410101001

83.33

105

110100189.29

360100100

83.33

92

1011100

89.29

⦙

⦙

83.33

⦙

⦙

89.29

37

0100101

84.03

⦙

⦙

89.29

48

0110000

83.33

112

1110000

89.29

380100110

84.75

108

110110089.29

490110001

84.03

113

1110001

90.09

⦙

⦙

84.75

109

1101101

90.09

⦙

⦙

84.03

⦙

⦙

90.09

41

0101001

84.75

⦙

⦙

90.09

56

0111000

84.03

120

1111000

90.09

127

111111190.09

570111001

84.75

⦙

⦙

84.75

64

1000000

84.75

Rz-p

Sz-p

Rz-p+1 ···· Rz-q

Sz-p+1 ····

···· Rz-1···· Sz-1

Sz-q

Rz

TABLE II SPECIFICATIONS OF PROTOTYPE.

LSB(1bit)

Sz

Coupling coefficient

k

0.20

Primary inductanceL1

392 H

Secondary inductanceL2 H

Primary capacitanceC1

8.96 nF

Secondary capacitanceC2

8.78 nF

Transmission distancel

150 mm

····

420 V

····

V in

····

Input DC voltage

····

Value

····

SymbolXz-p

Xz-p+1 ···· Xz-q···· Xz-1

Xz

MSB(7bit)

X z  X z  p  X z q
Fig. 5 Generation method of 7-bit pseudorandom numbers based on a
maximal length sequence.

C1
Vin

+

i1

v1

Fig. 6 Experimental setup.

SCH2080KEC (ROHM)

DiodesSCS220AE (ROHM)

Ferrite plates

PC40 (TDK)

C2

k
L1

MOSFETsR
L2

vo

C. Operation Waveforms.
Fig. 9 shows the operation waveforms. In all operation
methods, the output power is fixed at 3.0 kW. Fig. 9 (a) shows
the waveforms obtained using the conventional method with

Coil

Current direction
Probe
500

mm

500 mmWinding
z

z

y

y

x

x

Fig. 8 Radiation noise measurement setup.

SecondarywindingPrimary voltage v1 400 V/div
0

Ferriteplate
0Primary current i1 10 A/div

Primary
winding0

Fig. 7 Transmission coils with a rated power of 3-kW (Solenoid-typecoils).

the output frequency fixed at 85.1 kHz. Fig. 9 (b) and (c) show
the waveforms obtained using SSUD and SSBD, respectively.
The output frequency of the voltage source inverter in thesecases is selected randomly according to Table I (a) and (b),respectively. To obtain the results shown in Fig. 9 (b) and (c),the output frequency was varied from 80 to 90 kHz at fixed
time intervals according to the generated pseudorandom
numbers. When the proposed methods are used, the amplitudeof the primary current i1 varies. However, constant output
voltages were obtained for all operation methods.
D. Radiation Noise.
Figs. 10–11 show the radiation noise at point A in the x-y
and y-z planes defined in Fig. 8. Fig. 12 shows the radiationnoise at point B in the y-z plane defined in Fig. 8. In Figs. 10–
12, subpart (a) shows the results obtained using theconventional method, and subparts (b) and (c) show the results
obtained using SSUD and SSBD, respectively. When theconventional method was used, the fundamental and low-orderharmonic components sharply appeared. When the proposed
methods were used, the maximum value for the fundamental
and low-order harmonics were suppressed. When SSUD and
SSBD were used, the harmonic components around thefundamental frequency were suppressed by 4.45 and 8.27
dBA in comparison with the conventional method, which
operates the inverter at a fixed frequency. In the same manner,
the low-order harmonic components were suppressed incomparison with the conventional method. Both of the
proposed methods can be used to suppress these components.

Output voltage vo 400 V/div
10 s/div

Primary voltage v1 400 V/div
00Primary current i1 10 A/div0

Output voltage vo 400 V/div
10 s/div

Primary voltage v1 400 V/div
00Primary current i1 10 A/div0

Output voltage vo 400 V/div
10 s/div

Fig. 9 Operation waveforms of the IPT system. (a) Constant frequency.
(b) Proposed method I: SSUD. (c) Proposed method II: SSBD.

However, in the operation with SSUD, the harmoniccomponents peaked at 85.1 kHz. This was caused by thefrequency dependence of the impedance. In addition, theproposed method causes higher noise floor due to the change[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

-20
100

225

350

475

[kHz]

100

[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

225

350

475

[kHz]

-20
80.0

82.5

85.0

87.5

[kHz]

80.0

[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

82.5

85.0

87.5

[kHz]

-20
100

225

350

475

[kHz]

100

[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

225

350

475

[kHz]

-20
80.0

82.5

85.0

87.5

[kHz]

80.0

[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

82.5

85.0

87.5

[kHz]

-20
100

225

350

475

[kHz]

100

[dBA]
60[dBA]
60

40

40

20

20

0

0

-20

225

350

475

[kHz]

-20
80.0

82.5

85.0

87.5

[kHz]

80.0

82.5

85.0

87.5

[kHz]

Fig. 10 Radiation noise (Point A, x-y plane). (a) Constant frequency. (b)Proposed method I: SSUD. (c) Proposed method II: SSBD.

Fig. 11 Radiation noise (Point A, y-z plane). (a) Constant frequency. (b)Proposed method I: SSUD. (c) Proposed method II: SSBD.

of the output frequency. The proposed system should be
evaluated whether it satisfies the regulation. However, themeasurement method for the regulation (e. g. measurement
environment, equipment) are still in discussion. It is expectedthat the proposed system will be evaluated after the official
regulation is established.

maximum efficiency was 94.9% at an output power of 3.0 kW
when the inverter was operated at a fixed frequency. In
contrast, the maximum efficiency was 94.1% at an output
power of 3.0 kW when SSUD was used. The decrease in theefficiency is caused by the increased reactive current due tothe difference between the operating and resonancefrequencies. The reactive current increases the copper loss,
iron loss, conduction loss, and switching loss on the converter.
Hence, the decrease in efficiency with the SSUD was 0.8% at
the rated load. When SSBD was used, the maximum
efficiency was 93.8% at the rated load. The decrease in theefficiency was 1.1% in comparison with that of theE. Efficiency Evaluation.
Fig. 13 shows the DC-to-DC efficiency characteristics forthe conventional and two proposed methods. The efficiency isdefined as the ratio of the input DC power to the output DC
power. All of the curves show similar characteristics. The

100
100

Efficiency (DC-to-DC) [%][dBA]
60
40
20
0
-20
100

225

350

475

[kHz]

[dBA]
60
40

95
95

Constant frequency

Proposed method I(SSUD)

90
90

Proposed method II(SSBD)

85
85

20

80
80
1000
1000

0
-20
80.0

82.5

85.0

87.5

[kHz]

1500
1500

2000
2000

Output Power [W]

2500
2500

3000
3000

Fig. 13 Efficiency characteristics.

[dBA]
60

V. CONCLUSION

40
20
0
-20
100

225

350

475

[kHz]

[dBA]
60
40
20
0
-20
80.0

82.5

85.0

87.5

[kHz]

[dBA]
60
40
20
0
-20
100

225

350

475

[kHz]

[dBA]
60
40
20
0
-20
80.0

82.5

85.0

87.5

[kHz]

Fig. 12 Radiation noise (Point B, y-z plane). (a) Constant frequency. (b)Proposed method I: SSUD. (c) Proposed method II: SSBD.

conventional method. Thus, using SSBD is effective in theheavy-load region. In contrast, the efficiency drops by amaximum of 4.4% when the output power is 1.0 kW. In thelight-load region, the effect of the no-load loss is larger thanthat in the heavy-load region. However, the current in thewindings is smaller than that in the heavy-load region.
Considering that the radiation noise is proportional to thecurrent, weak suppression of the radiation noise can be
acceptable in the light-load region. Therefore, in the light-loadregion, SSUD should be used.

In this paper, two methods for the reduction of the radiationnoise of an inductive power transfer system were proposedand experimentally demonstrated. The radiation noise fromthe transmission coils for the IPT system was spread in thefrequency domain by changing the output frequency of thevoltage source inverter at random. This eliminates thenecessary of additional components, such as a noise shield anda filter circuit, in the proposed methods. In the first proposed
method, called SSUD, the output frequency of the voltage
source inverter is selected from the range of 80 to 90 kHz
from a discrete uniform probability distribution. In the second
proposed method, called SSBD, the output frequency is
selected from a biased discrete probability distribution. The
probability distribution is proportional to the combined
impedance from the transmission coil and the compensation
capacitor on the primary side considering the frequency
characteristics of the IPT system. Because of this bias, thefrequency components of the output current, which flows
through the transmission coils, was from 80 to 90 kHz. Fromthe experimental results with the 3.0 kW prototype, theharmonic components around the fundamental frequency ofradiation noise were suppressed by 4.45 and 8.27 dBA by
applying SSUD and SSBD, respectively. Therefore, SSBD
was more effective than SSUD and the conventional methodin the suppression of the noise expect in the heavy-load region.
In the light-load region, using SSBD can be greatly decrease
the DC-to-DC efficiency because of the no-load loss, which
dominates the loss. Thus, SSUD is a more suitable reduction
method in the light-load region because SSUD provides
radiation noise reduction with a small decrease in theefficiency in comparison with the conventional method.
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Kent Inoue (S’16) was born in Ibaraki, Japan, in 1993.
He received his B.S. degrees in electrical, electronics
and information engineering from Nagaoka Universityof Technology, Niigata, Japan in 2016. He has beenwith Nagaoka University of Technology as a M.S.
course student in electrical, electronics andinformation engineering. His current research interests
includes an inductive power transfer system.
Mr. Inoue is a student member of the Institute ofElectrical Engineers of Japan and the IEEE.
Keisuke Kusaka (S'13, M'16) was born in Miyagi,
Japan, in 1989. He received his B.S. and M.S. degrees
in electrical, electronics and information engineeringfrom Nagaoka University of Technology, Niigata,Japan in 2011, 2013, respectively. From 2015 to 2016,he was with Swiss Federal Institute of Technology in
Lausanne (EPFL), Switzerland as a trainee. In 2016,
he received his Ph.D. degree in energy andenvironment science from Nagaoka University ofTechnology. Since 2016, he has been with Nagaoka
University of Technology, Niigata, Japan as a researcher. His current researchinterests include the areas of inductive power transfer systems and highfrequency converters.
Dr. Kusaka is a member of the Institute of Electrical Engineers of Japan,
the Society of Automotive Engineers of Japan and the IEEE.

Jun-ichi Itoh (M’04, SM’13) was born in Tokyo,
Japan, in 1972. He received his M.S. and Ph.D.
degree in electrical and electronic systems
engineering from Nagaoka University of Technology,
Niigata, Japan in 1996, 2000, respectively. From1996 to 2004, he was with Fuji Electric Corporate
Research and Development Ltd., Tokyo, Japan. He
was with Nagaoka University of Technology, Niigata,Japan as an associate professor. Since 2017, he has
been a professor. His research interests are matrix
converters, dc/dc converters, power factor correction techniques, energy
storage system and adjustable speed drive systems.
He received IEEJ Academic Promotion Award (IEEJ TechnicalDevelopment Award) in 2007. In addition, he also received Isao Takahashi
Power Electronics Award in IPEC-Sapporo 2010 from IEEJ, 58th OHM
Technology Award from The Foundation for Electrical Science andEngineering, November, 2011, Intelligent Cosmos Award from Intelligent
Cosmos Foundation for the Promotion of Science, May, 2012, and Third prize
award from Energy Conversion Congress and Exposition-Asia, June, 2013.
Prizes for Science and Technology (Development Category) from theCommendation for Science and Technology by the Minister of Education,
Culture, Sports, Science and Technology, April 2017.
Dr. Itoh is a senior member of the Institute of Electrical Engineers ofJapan, the Society of Automotive Engineers of Japan and the IEEE.