A Novel Current Ripple Cancellation PWM
for Isolated Three-phase Matrix DAB AC-DC Matrix Converter
Shunsuke Takuma, Keisuke Kusaka, Jun-ichi Itoh , *Yoshiya Ohnuma, *Satoshi Miyawaki
Nagaoka University of Technology/ *Nagaoka Power Electronics co., Ltd.
1603-1 Kamitomioka-chi Nagaoka city Niigata, Japan
Tel.,Fax: +81 / (258) – 47.9533.
E-Mail: takuma_s@stn.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp
URL http://itohserver01.nagaokaut.ac.jp/itohlab/index.html

Keywords«Matrix converter», «Three-phase system», «Modulation strategy», «Power factor correction»,«Battery charger»

Abstract
The control method for an isolated three-phase matrix dual active bridge (DAB) AC to DC converter
is proposed to control sinusoidal grid current waveforms with a minimum reactive current. The
conventional SVM based on a constant current is not applied to the isolated three-phase matrix DAB
AC to DC converter due to a high di/dt by employing a leakage inductance. Therefore, the duty
calculation for the matrix converter is required to approximate the waveforms at a high-frequency
transformer or implement a numerical calculation. However, the gird currents are distorted due to theerror by such approximation. In addition, the numerical calculation is used as a complex implementation
and requires a long calculation time. In this paper, the novel modulation method based on the current
ripple cancelation is proposed for the isolated three-phase matrix DAB AC to DC converter. The highdi/dt ripple current against current reference is canceled by the sum of the currents during the positive
and negative half switching cycle without the leakage inductance value. The input and output dc current
controls are achieved considering the effects of the leakage inductance. A continuous current mode
(CCM) and discontinuous current mode (DCM) on buck and boost modes are clarified and confirmedby simulation. As the simulation results, the input current THD is reduced by 88.2% compared with theconventional method. As experimental results, the input current THD of 3.8% is achieved at rated powerwith CCM and DCM operation.

Introduction
In recent years, a demand for a rapid charger from a three-phase grid to a battery with galvanic
isolation is increased. Requirements of a rapid charger are a constant output DC voltage, three-phasepower factor correction (PFC) operation, and galvanic isolation between three-phase grid and batteryside. In addition, downsizing and a weight reduction are also required. Generally, the isolated threephase AC to DC converter employs a medium or high-frequency AC transformer to reduce the volumeof the transformer. However, grid-tied inductors and smoothing capacitors maintaining a constant DC
voltage prevent the downsizing of the system. Therefore, the isolated AC-DC converter using the matrix
converter at the primary side has been studied [1]-[4]. A high-frequency voltage from a grid voltage isdirectly generated by the matrix converter. The downsizing and high efficiency are expected due to theelimination of the passive components such as e.g. the boost inductors and the DC-link capacitors.
In order to minimize further the system, the leakage inductance of the transformer is utilized such a
DAB converter [5]-[6]. The control method for the isolated single-phase matrix DAB AC to DC
converter is proposed in [7]-[8] based on the DAB control theory. However, the single-phase system is
not suitable for the large capability application due to a high conduction loss compared with the threephase system. In such isolated three-phase matrix DAB AC to DC converter, the control of the three-Input filter

phase to single-phase matrix converter at the primary side and the full bridge converter at the secondaryside is required. Generally, the control of the isolated three-phase matrix DAB AC to DC converter isdeveloped from the control theory for the dual active bridge (DAB) converter, which is based on theassumption that the input and output voltages are constant during one switching cycle. However, theinput voltage of the isolated three-phase matrix DAB AC to DC converter changes according to the
input voltage, which is the grid voltage. In addition, the high-frequency current has a high di/dt due to aleakage inductance of the transformer, leading to a nonlinear relationship between the duty and thetransmission power. Therefore, the complex control algorithm is required to control the transmission
power and the sinusoidal waveform at the input side. In order to simplify the duty calculation, theapproximation of the output voltage of the matrix converter by the square waveform is proposed [9].
However, the input current distorts due to the error between the approximation waveform and the actual
waveform. Meanwhile, the duty calculation with numerical analysis is proposed [10], which still results
in a complex calculation. In [11]-[12], a simple control algorithm, which is the phase-shift control
between the primary voltage and the secondary voltage has been proposed. However, the circulating
current in the matrix converter increases compared with the control method of [10], resulting in the
conduction loss increase.
S1
S3
Srp
Ssp
Stp
In this paper, the control method for thePrimary Secondry
isolated three-phase matrix DAB AC to DC
iout
converter is proposed to achieve the low
ir
vr
is
vs
L
input current distortion without a numerical
vpr
vse
it
vt
calculation. The original idea of the
proposed control method is that the AC
current reference and periodic average
Srn
Ssn
Stn
S2 S4
current at the transformer are matched by
3f -1f matrix converter
PWM
rectifierthe ripple cancellation of the active current
during one switching period. The Fig. 1. High-frequency link matrix converter. This converter
cancelation method is achieved by the has a high di/dt because only small leakage inductor uses at
the high-frequency transformer.
selection of suitable output vector which is
employed by the conventional SVM. In
vb
010
010
V
V 001
addition, the output vector is selected with100
the consideration of the minimization of the010
Sector II
V 001
Sector I
reactive current. It is confirmed that the low
I
Sector III
I
100
input current distortion is achieved by the 001 V
Sub
Sector I
V 001
I
'
V
sector B
100
Iva
proposed control method.
Sub
3

2

2

2

*

in

4

1

1

z

Sector IV

Principle of current ripple
cancellation
Figure 1 shows the configuration of theisolated three-phase matrix DAB AC to DC
converter. The full bridge converter at the
primary side of a DAB converter is replacedby the matrix converter. In the control
method for this topology, the leakage
inductanceofthehigh-frequency
transformer is utilized. Consequently, thedownsizing and the high efficiency of thesystem are achieved by removing both thesmoothing inductor at the secondary sideand the snubber circuit.
Figure 2(a) shows the fundamental
voltage vector diagram and the reference
current vector Iin*. If the reference current
vector Iin* is generated by I1 and I2, the input

1

sector A

Sector VI

I2'

V1

Vz

Sector V

100
001

Vz : Zero vectors

001 V
5
010

V6

※The number of ”1”means
switch is Turn onEx.) V1

100010
100 Srp = ON, Ssp = OFF, Stp = OFF
001 Srn = OFF, Ssn = OFF, Stn = ON(a) Fundamental voltage vector diagram
200

vu

Phase
voltage 0vvvw[V] 200
VA

Sector

IA

IB

IIA

IIB

IIIA IIIBIVA IVB

x

v2

100

I1

Current 20
reference[A]

VIA VIB

v1

Line 300
voltage 200
[V]

VB

0

I2
0

5

10

15

20 [ms](b) Relationship between grid voltage and sector
Fig. 2. Fundamental space vector modulation.

current becomes sinusoidal when output voltage vectors V1 and V2 are selected.
v2 b ia  v2a ib
I1 v1a v2 b  v1b v2a
I2 

(1)

v1a ib  v1b ia(2)v1a v2 b  v1b v2awhere v1a, v2a, ia and v1b, v2b, ib are alpha and beta components of V1, V2, and Iin*. In this paper, thecontrol method based on sector IA is explained. The other sectors operation are similar because thegrid voltage relationships is symmetric in every 30 degree.
Figure 2(b) shows the relationship between the grid voltages and the selected sectors and vectors.
Note that sectors and sub sectors are defined as Fig. 2(b) based on the grid voltages. The high-frequency
output voltage vector is generated according to the sectors and the sub sectors.
Conventional SVM
Figure 3(a) shows the high-frequency waveforms at the transformer with the conventional SVM. It is
assumed that two amplitudes at point A and B are equal to those at point A’ and B’, respectively. The
output voltage vectors V1 and V2 are selected to output the positive voltage during the first half cycle.
On the other hand, the output voltage vectors V4 and V5 are selected to output the negative voltage during
the last half cycle. The solid line and dot line depict the instantaneous currents from point A to B and A’
to B’ during the transition of the output vectors V1, V2 and V4, V5 respectively. The instantaneous currents
from point A to B and A’ to B’ during the positive and negative half cycle are same. As a result, theerrors between the current reference and each actual average current occur during the duration T1 and T2
vpr

V1

v1
v2

V2

Vz

0

V4

V5

p

-v2
-v1

I2
2T2

2p

i2

I1
2T1

vse
NVdc

Iave

0

p

i0
B

A

iu

A’
A

iv

0

p

A’
T1

B

Error2Error1

2p

-NVdc

ipr

B’

T2

Tz

iu

iv

T1

T2

2p

0

T1

T2

B’

(a) Convectional method
vpr

V1

v1
v2

V2

Vz

0

V5

V4

p

-v2
-v1

2p

vse

B’

B
I1
2T1

i2

NVdc
0

p

2p

I2
2T2

Iave

-NVdc

ipr

i0 A

B

A

iu

iv

0

p

T1

A’

T2

A’
Tz

iv

iu

2p

0

T1

T2

B’
T2

T1

(b) Proposed current ripple cancellation method
Fig. 3. Effect of current ripple. The error between current reference and actual current occurs due to the current
ripple with the conventional method. The current ripple during the positive cycle is canceled by them during thenegative cycle with the proposed method.

because the ripple current is not ignored due to leakage inductance. Therefore, the conventional SVM
is not suitable for the isolated three-phase matrix DAB AC to DC converter because the conventional
SVM is based on constant output current during switching periods. In order to solve this problem, acomplex algorithm or an offline calculation is required to include the effect of the current ripple. In thispaper, the current ripple cancellation method is proposed to keep constant output current regardless ofthe small leakage inductance.
Proposed SVM for current ripple cancellation
Figure 3(b) depicts the ripple cancellation principle for the active current during one switching period.
The instantaneous currents from point A to B and A’ to B’ during the positive and negative half cycle
are different because the primary voltage during the negative cycle is changed to cancel the current
ripple. The average currents Iave1, Iave2 during durations T1, T2 are expressed asv pr1  NVdc
v pr1  NVdc 1
I ave1  i0 
T1  i2 
T1 
(3)
2
L
L

I ave 2 v pr 2  NVdc
v pr 2  NVdc 1T2  i2 
T2 
i0 
2
L
L


(4)

i i
I ave  I ave1  I ave 2  0 2
2(5)
where, v1 and v2 are the output voltages during V1 and V2 respectively, i0 is the initial current at V1, andi2 is the final state at V2. The average current is independent of the high di/dt due to the leakage
inductance. Therefore, the linearization between the input current reference and the duty are achieved
to cancel the active current ripple during one switching periods. The primary and secondary voltages
are generated by the matrix converter and the inverter based on the principle of the cancellation method.

Control Strategy

Switching signal

In this section, the details of the proposed control method based on SVM and the switching pattern
are explained.
Figure 4 shows the switching pattern for the matrix converter, the PWM rectifier and the operation
waveforms at the transformer.
 t0-t1 The zero voltage at the primary side is generated by the short circuit operation between
the upper and lower switches in one leg of the matrix converter, i.e.. S2 and S3 at the PWM
rectifier turn on to clamp zero voltage
 t1-t2 The matrix converter outputs the maximum line to line voltage v1 at the grid voltages. The
secondary voltage is the negative voltage to minimize the reactive current. Next switching
starts when the output current reaches to point A.
 t2-t3 The PWM rectifier outputs the
Matrix P T
positive voltage. This duty is decided
R
S
T
R T
converter N
T
R
T
R S
R
T
by the current reference based on
SVM.
S
PWM S
 t3-t4 Srp and Ssp of the matrix
rectifier S
S
converter turns off and turns on. The
power is transmitted from the gridside to DC side during t2 to t4.
vpr 0
 t4-t6 The maximum line to line
voltage is selected to minimize thevse 0
reactive current. Therefore, Ssp andB
Stn of the matrix converter turn off.
I
AC
Stp and Srn of the matrix converter
iout 0O
O
turn on.
B’ C’A’ t6-t7 Both voltage at the primary and
t
t t
t
t t t
t t
t
t t t
secondary side is maintained to zeroFig. 4. Ripple cancellation principle for active current.
to clamp zero current.
1
2
3
4

ave

0

1

2

3

4 5

6

7

8

9

10 11 12t7-t8 The matrix converter outputs the maximum line to line voltage v1 at the grid voltages. The
secondary voltage is the negative voltage to minimize the reactive current. Next switching
starts when the output current reaches to point A’ which has the same amplitude of point A.
 t8-t9 the matrix converter outputs the medium line to line voltage v2 to cancel the current
ripple.
 t9-t10 Ssn and Srn of the matrix converter turns off and turns on. ZVS when Srn turns on does notachieve because the drain-source voltage of Srn cannot be discharged due to the negative
direction. However, the drain-source voltage of Srn is small voltage which is the differential
voltage between the maximum and medium line to line voltage.
 t10-t12 In order to minimize the reactive current, the maximum line to line voltage is output
from the matrix converter. Therefore, Stp and Srn of the matrix converter turn off. Srp and Stn ofthe matrix converter turn on.
Figure 5(a) and (b) show a high frequency current waveform with the current ripple cancellation. The
operation modes are separated into two modes such as buck mode and boost mode depending on thegrid voltages and DC voltage. The sign function is defined as
 1 x  NVdc
v I v Isign( x)  x 1 1 2 2(6).
I1  I 2
 1 x  NVdc
The average current Iave during point A to B is defined as the equivalent transformer current reference
Iave* in the powering mode. Each duty of I1 and I2 expressed asID1  *1  sign ( x ) Dz
(7),
I ave
I2
(8).
*
I ave
where Dz is the compensation duty for D1. The current during Tz is a reactive or active current in thebuck or boost mode. Therefore, addition or subtraction from D1 is required to compensate the effect of
the current during Tz. As shown in Figure 4(b), the sum of the output current, which is from point 0 to
A and from point 0 to B, is double of Iave* and is expressed as(v  NVdc )(v  NVdc )(v  NVdc )1*
2 1
T ( Da  Dz (1  sign( x)))  1
TD1  2TD2  2 I ave
(9)
L
2
L
L
where L is the leakage inductance referred to the primary side of the transformer, Vdc is the DC voltage.
Solving the equation (8), Da is expressed as
L
2 I ave*  (v1  NVdc ) D1  (v2  NVdc ) D2
1
TDa  Dz (1  sign( x ))(10).
2(v1  NVdc )2
D2 Active mode Circulating mode
Active mode

v1
v2
vpr 0
-v2
-v1
NVdc
vse 0
-NVdc
Iave

iout 0Circulating mode

Active mode

-v2
-v1
NVdc
vse 0
-NVdc

B

B

Iave

A
C

iout

O

O

B’ C’

A’
T0 Ta

T1

T2 T z T a

T0

Ta T2

T1

Active mode

v1
v2
vpr 0

Tz Ta T0A

C
0O
T0 Ta Tz

O
C’ B’
T1

T2 Ta

T0

Ta Tz T2

A’
T1(a) Buck mode v1>NVdc>v2(b) Boost mode NVdc >v1 >v2
Fig. 5. Ripple cancellation principle for active current.

Ta T0

Dz is decided to equalize the amount of the change during T1 to T2 respectively.
(v  NVdc ) I1  (v2  NVdc ) I 2
Dz  sign ( x ) 1(11)*
2 NVdc I ave
Substring (11) into (6), D1 is expressed by I (V2  Vout ) D2  (V1  Vout )D1   *1(12)
 I ave V1  Vout  2Vout
Finally, the zero current period D0 is defined by(13)
D0  1  (2Da  D1  D2  Dz )*The equivalent transformer current reference Iave is restricted by the switching frequency, the leakage
inductance L, the grid voltage, and the DC voltage. The equivalent transformer current reference isexpressed as2
v1  NVdc T1  v1  NVdc T  v1  NVdc I1 (v1  NVdc )  I 2 (v2  NVdc ) T *
I ave  (1  D0 )
 


2
2
L
2
L 
2
NVdc
L



(14)
In CCM, the zero current period is keep to zero. Therefore, the current reference for CCM is expressedas2
1  v1  NVdc T v1  NVdc T  v1  NVdc I1 (v1  NVdc )  I 2 (v2  NVdc ) T *
I ave  
 

(15)


2
2
L
2
L
2
NVdc
L



On the other hands, the high frequency current has zero current periods in the DCM which is operated
under Da=0. Therefore, the zero current period D0 is expressed asI I
D0  1  1 * 2
(16)
I ave
The current reference for the DCM is expressed asv  NVdc
T
I ave*  sign( x ) 1((v1  NVdc ) I1  (v2  NVdc ) I 2 )(17)
4 NVdc
L
The current references for CCM and DCM are same at the boundary condition. Substituting (17) into
(15), the transition power at the boundary condition is expressed asp NVdc 3vac 2  2 N 2Vdc 2(18),
Pb _ buck L
4 6vac1.4

Max. f(x)Boost mode

1.0

Buck boost mode

0.8
0.6

DCM

Min. f(x)Buck mode

1.2

NVdc / 2 vac

1.2

NVdc / 2 vac

Eq. (19)

1.4

CCM+DCM

CCM

0.8
0.6

DCM

0.4

0.4

Maximum power
Eq. (20)1

Eq. (18)%Z=0.05%

0.2

Not operate

0.2

1p.u. = 3 kW

0

0

1

Power[p.u.]

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Power[p.u.](a) Operation modes(b) Boundary conditions
Fig. 6. Relationship between maximum transition power and input and output voltage. The operation modeand current mode are changed by the grid voltage, DC voltage and transition power.

p vacN 2Vdc 2  2vac 2(19)
L
2 2 NVdc
where vac is the RMS value of the grid voltage.
The equations (18) and (19) express theboundary condition without CCM during 0 to
30 degree completely. Solving (14), themaximum transition power P is expressed as
p 6 vac NVdc
P
(20)
16
L
Changing (20), the leakage inductance iscalculated by using (21).
p 6 vac NVdc
L
(21)
16
P
Figure 6(a) shows the operation mode of theisolated three-phase matrix DAB AC to DC
converter. It is assumed that the turn ratio of thetransformer is 1. The matrix converter isoperated under the boost mode when the DC
voltage is larger than the maximum line to line
voltage. On the other hands, the matrix
converter is operated under the buck mode
when the DC voltage is lower than 0.866 timesof the maximum line to line voltage.
Figure 6(b) shows the boundary conditions
between CCM, DCM and mixed CCM and
DCM during the grid period. %Z of the leakage
inductance is designed to 0.05%. According to(20), the transition power is restricted by theswitching frequency, the leakage inductance,
the grid voltage, the DC voltage, and the turnratio. A DAB converter for the DC-DC
conversion has the similar limitation by thesame circuit parameter. The high-frequency
current has zero current period in DCM when
the DC voltage is significantly larger or lower
than the grid voltage at a light load.
Pb _ boost 

Table I Simulation and experimental conditions
Element

SymbolValue

Rated power
Three-phase AC voltage
DC voltage
Input frequency
Carrier frequency
Leakage inductanceTurn ratio of transformer
Input filter
Dead-time

P
vac
Vdc
f
fsw
L(%Z)
N1:N2
Lf
Cf
Td

3.0 kW
200 V
120 V
50 Hz
20 kHz45 µH(0.2%)
2:1
1 mH
10 mF
250 nsGrid voltage[V]

200
0
-200

U-phase current[V]

10
0
-10
Periodic average of 2T
Primary voltage[V]

200
0
-200
Secondary voltage[V]

100
0
-100
Primary current[A]

10
0
-10
1.0

0.0

CCM DCM CCM DCM CCM DCM CCM DCM CCM DCM

D1
D0
D2

0

Fig. 7(b)20 [ms]Fig. 7(c)(a) operation waveforms
Primary voltage[V]

200
0
-200
Secondary voltage[V]

100
0
-100
Primary current[A]Simulation and Experimental
Results
Table 1 shows the simulation andexperimental conditions. The four-step voltage
commutation is applied to avoid the shortcircuit for the matrix converter. Each dead-time
for the matrix converter and the PWM rectifier
is set to 250 ns depending on the rise and falltime of the switching device (ROHM
SCT2080KE).
Figure 7 shows the input voltages andcurrents of the matrix converter and the
waveforms at the transformer without thecommutation and the input filter to confirm thevalidity of the proposed control method at light

10
0
-10
T0

T1

T2

T0

T2

T1

5 ms

T2

5 ms

(b) DCMPrimary voltage[V]

200
0
-200
Secondary voltage[V]

100
0
-100
Primary current[A]

10
0
-10
T1

T1

T2

(c) CCM
Fig. 7. High-frequency voltage and current waveforms at
transformer.

load (0.2 p.u.). The operation mode is changed by (6) from CCM to DCM during 6 times of the gridfrequency. It is confirmed that the input current is not distorted by changing two mode because each
duty can be calculated at the peak or bottom of the carrier. The input current is controlled to thesinusoidal waveforms and unity power factor. Fig. 7(b) and (c) shows the high frequency waveforms at
the transformer with CCM and DCM. The each operation mode is changed by the relationship between
the instantiations line to line voltage and DC voltage. In DCM, the primary and secondary voltage iscontrolled to keep the zero voltage when the current at the transformer is zero. On the other hand, thecurrent at the transformer is operated in CCM.
Figure 8(a) and 8(b) show the simulation results of the actual and reference phase-current with theconventional method [10] and the proposed method, respectively. The input current filter uses theperiodic average filter by the switching frequency without the LC filter to confirm each theory. In Fig.
8(a), the actual input current follows the command only when the approximated square waveforms andactual waveforms are matched. However, the input current distorts due to the error which is not ignored
when the approximated square waveforms and actual waveforms are different. In Fig. 8(b), the actual
current completely follows the command with the proposed current ripple cancelation method. It is
confirmed that the grid current is controlled to keep the sinusoidal waveforms by the proposed method.
Figure 9 shows the comparison results of THD characteristics between the conventional and proposed
method with the LC filter. The input current THD characteristics of the conventional method and the
proposed method. The input current THD with the conventional method is over than 15.0% under the
any load condition due to the approximation error. The THD requirement for the grid connection is notsatisfied at rated power. It is difficult to improve THD by a compensation such as feed-forward because
the error between the actual current and current reference is nonlinear. On the other hand, the input
current distortion with the proposed method is greatly reduced by 88.2% from 17.0% to 2.0% comparedwith the conventional method [10]. In addition, the duty for each converter is decided by the equation
without a numerical calculation and off-line calculation. The input current THD which is lower than 5%
is achieved over the load range from 0.3p.u. to 1.0p.u.

Acurual current ir

I

Acurual current ir

Current reference ir*
0

2p

p

R-phase current

R-phase current

I

-I

-I

Grid frequency 50 Hz

Current reference ir*

0

2p

p

Grid frequency 50 Hz

Conventional control method
Proposed control method
Fig. 8. Simulation results of actual value and command. The input filter is the periodic filter of theswitching frequency. The input current reference and the actual current are completely matched with theapplication of the proposed method.
30

With conventional method

1p.u. = 3 kW
%Z = 0.2%

Input current THD[%]

25
20
15

Reduced by
88.2%

With proposed method

10
5
0
0

0.2

0.4

0.6

0.8

1

Power[p.u.]

Fig. 9. THD characteristics without commutation. The input current THD is suppressed by 88.2% to apply the
proposed method. The input current is controlled to keep the low distortion which is lower than 5% over the
input power from 0.3p.u. to 1.0p.u..

Figure 10(a) shows the experimental results of the input and output waveforms of the matrix converter
at rated power. The input current THD of 3.8% and the unity power factor are achieved by theapplication of the proposed method, satisfying the grid harmonic constraints. Figure 11(b) shows theextended waveforms of Fig. 11(a). It is confirmed that the output voltage waveforms are similar between
the experimental and simulation results.
Figure 11(a) and (b) show the primary side voltage, secondary side voltage and primary side current
at the transformer in DCM and CCM at the light load. In DCM, the primary current is maintained tozero current when the primary voltage and secondary voltage is zero voltage in theory. However, theprimary current has the offset of 2A due to the effect of the dead-time. The actual switching timing isdelayed by the dead-time depending on the direction of the high-frequency output current. The deadtime compensation will be implemented to improve the input current THD. In Fig. 11(b), the output
current is controlled to remove zero current period to minimize the reactive current. Therefore, the zerovoltage period is also removed from the primary voltage and secondary voltage for CCM

Conclusion
The control method for the isolated three-phase matrix DAB AC to DC converter, which consisted ofthe matrix converter and the PWM rectifier was proposed. The current ripple during one switching
period was cancelled by the sum of the ripple current during the positive and negative half switchingcycle without the leakage inductance value. Therefore, the current ripple cancellation method was not
required to calculate the complex duty because the duty was calculated by SVM based on constant
output current. The active current was controlled with the ripple cancellation due to the leakage
inductance of the transformer. The boundary conditions between CCM and DCM on the buck and boost
mode were employing by the improvement of the efficiency with a pulse frequency modulation at lightload. By applying the proposed method, the input current THD at rated load was suppressed by 88.2%

vr 200[V/div]
vr 200[V/div]00
ir 20[A/div]
ir 20[A/div]00
vpr 300[V/div]

vpr 300[V/div]00iout 20[A/div]00iout 20[A/div]

Fig. 10(b)10[ms/div]

5[ms/div](a) Operation waveforms(b) Extended waveforms of Fig. 10(a)Fig. 10. Experimental results of input and output waveforms. The input current THD of 3.8% is achieved by the
proposed method.
10[ms/div]

10[ms/div]

Active Reactive Active Reactive
current current current current

vpr 200 V/div0

vpr 200 V/div0
vse 100 V/div0

vse 100 V/div0
ipr 20A/div

ipr 20A/div

0

Active Reactive Active Reactive
current current current current

0(a) Discontinues current mode(b) Continues current mode
Fig. 11. Operation waveforms at high frequency transformer. (P=0.2 p.u.)in the simulation. As experimental results, the input current THD is 3.8% at rated power in CCM and
DCM.

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