Surge Voltage Reduction Method
for DAB Matrix Converter
using Circulating Current in Whole Load Condition
Shunsuke Takuma
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
takuma_s@stn.nagaokaut.ac.jp

Kashin Kiri
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japans101622@stn.nagaokaut.ac.jp

Hiroki Watanabe
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japanhwatanabe@vos.nagaokaut.ac.jp

Jun-ichi Itoh
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
itoh@vos.nagaokaut.ac.jp
Abstract— This paper proposes a surge voltage reduction
method by achieving zero-voltage switching (ZVS) for a matrix
converter or an inverter of a dual active bridge (DAB) matrix
converter with circulating current in discontinuous current
mode (DCM). In traditional DAB converter, ZVS cannot be
achieved including a light load condition. The duty ratio
calculation for a circulating current is revealed to achieve an
input power factor correction and ZVS. From the experimental
results with a prototype, the ZVS operation is achieved byincreasing the circulating current even zero-transmission
power. In addition, the grid current distortion is lower than
5.0% at the rated power regardless of the DC voltage variationof plus or minus 20%.
Keywords— Three phase AC-DC converter, Matrix converter,
Discontinuous current mode

I. INTRODUCTION
In recent years, to increase the driving range of electricvehicles, the battery capacity has been increasing. An increasein the number of quick chargers installed on the ground isrequired. Isolated three-phase AC-DC converters are suitablefor high-capacity circuit configurations such as fast battery
chargers. This power converter has the capability of the power
factor correction (PFC) operation and galvanic isolation
between three-phase power grid and battery side. In addition,
the isolated three-phase AC to DC converter is required toregulate DC voltage of the battery because the battery voltage
is changed against nominal DC voltage depending on thebattery condition. Generally, the isolated three-phase AC to
DC converter using PWM rectifier employs a mediumfrequency AC transformer in order to reduce the volume of
the transformer. However, the grid-tied inductors and the DClink capacitors limits the miniaturization of the system
volume. On the other hand, the isolated AC-DC converter
using a matrix converter at the grid side has been studied
actively [1]-[6]. The matrix converter directly generates themedium-frequency voltage from the grid frequency voltage
without a DC stage. Therefore, the bulky grid-tied inductors
and DC-link capacitors are not necessary.
It has been proposed that the isolated AC-DC converter
using the matrix converter works on the same principle as dual

active bridge (DAB) converter [7]. The conventional control
methods for the DAB converter [8]-[11] are not applied to the
DAB matrix converter because the grid current must be
controlled sinusoidally according to the grid voltage.
Various soft-switching methods for the DAB matrix
converter have been proposed [12]-[18]. The soft switching
technique is achieved by combining the switching sequencewith the direction of the transformer current. The duty ratio is
calculated by approximating the transformer current
waveform to simplify the square waveform [16]. However, thelarge distortion occurs in the grid current due to theapproximation error. The ZVS operation in the DAB matrix
converter is achieved by completely discharging the parasitic
capacitance charge of the switching device [16]. Therefore,
the ZVS range is determined by the current direction, theparasitic capacitance, the dead-time period, and theinstantaneous current during switching. ZVS is not achievedin the light load region because the instantaneous current of
the switching device is not enough to the ZVS condition. In
the conventional method, the surge voltage occurs in thematrix converter side or the inverter side in the DAB matrix
converter. Therefore, the operating range is limited by thesurge voltage when the margin of the rated voltage of thepower device is small. On the other hand, the on-resistance ofthe switching device increases when the high rated voltage isapplied to the DAB matrix converter.
The modulation method that calculates the duty ratio
without the approximation has been proposed [17]-[18]. In
this method, the suppression of the grid current and zerocurrent switching (ZCS), and zero voltage switching (ZVS)
are achieved by the circulating current in a light load [18].
However, the conduction loss increases at a heavy load due tothe large circulating current.
This paper proposes the surge voltage reduction method ineither the matrix converter side or inverter side by achieving
ZVS in the whole load region. The original idea of this paper
is that the circulating current is optimized in order to achieve
ZVS regardless of AC and DC voltage amplitudes and atransition power condition. The proposed method achieves
ZVS by adjusting only the minimum circulating current in thewhole load condition. The ZVS operation can be selected fromeither the matrix converter or the inverter by the direction of
the circulating current. In addition, the grid current distortion
is suppressed by deriving the duty ratio considering theincrease of the circulating current. From experimental results,
the surge voltage is suppressed to less than 1.1 times of the
rated DC voltage by the proposed method.

vr ir Lac Cac
vs is
vt it

II. CIRCUIT CONFIGURATION AND CONTROL METHOD

Sps

Spt

Srp

Ssp

Stp

itr
L
vmc

Srn

Ssn

Stn

Snr

Sns

Snt

Cdc Vdc

vinv
N1:N2
S2

A. Without Circulating Current
Fig. 3 shows the output voltage of the matrix converter
and the output voltage and transformer current waveforms of
the inverter. The duty cycle is defined as Da, D1, D2, Db andzero voltage period D0, respectively. The zero-voltage period
D0 means that both the matrix converter and the inverter
output the zero voltage. The Da and D1 periods output the
maximum value v1 of line-to-line voltages. The D2 period
outputs the intermediate value v2 of line-to-line voltages. andthe Db period outputs the zero vector of the matrix converter.
Therefore, the transformer current during Tb does notcontribute to the transmission power. The instantaneoustransformer currents i1, i2, and i3 are expressed by (1).

S4InverterFig. 1. Circuit configuration of isolated three-phase AC to DC converter
using matrix converter.
vβ

010
V
100 3

010
V2 001

Sector I

I2

Sector III
001
V
100 4

I1'

Vz

Sector IV

010
V2 001

Vz : Zero vectors

Sector II

Iin * 100
V1 001

I1

Sub
sector BvαSector I

Sub
sector A

Sector VI

I2'

V1

Vz

Sector V
001 V
5
010

100
V6
010

100
001

※The number of ”1”means switch i s Turn on
100 Srp = ON, Ssp = OFF, St p = OFFEx.)
001 Srn = OFF, Ssn = OFF, St n = ON(a) Space vector diagram

Phase 200
voltage 0
[V]

vu

VA

VB

VIA VIB
IA

vw

IB

IIA

IIB

III A III B

IVA IVB

v1

Line 300
voltage 200
[V]

vv

-200

Sector

v2

100
0

I1

Current 20
reference

I2

0

[A]

0

0.5

1.0

2.0 [ms]

1.5(b) Relationship between sector and reference
Fig. 2. Space vector modulation for matrix converter.

MC

Matrix converter voltage vmc Swit ching signal
Inve rter volta ge Nv inv

Fig. 2 shows relationship between the medium-frequency
operation waveforms and the switching signals of the DAB
matrix converter. The circulating current assumes zero whenthe matrix converter and inverter output the zero voltage.
Therefore, the parasitic capacitance of the switching device is
not discharged when the upper arm of the matrix converter isturned from T-phase to R-phase. The ZVS method for thematrix converter side is that the circulating current i0 is
required to be the negative direction. On the other hand, theinstantaneous current is zero when S1 turns-on after S2 turnsoff in the inverter side. The ZVS operation is not achieved tokeep the charge of the drain to source voltage due to zerocurrent. To achieve ZVS in the inverter side, the circulating
current i0 is required to be the positive direction. In otherwords, ZVS is achieved only for either the inverter side or thematrix converter side based on the direction of the circulating
current i0.

S3S1

3φ -1φ matrix converter

Fig. 1 shows a circuit consisting of a three-phase to singlephase matrix converter, an inverter, and a medium-frequency
transformer. The transition power is decided by phasedeference between the output voltage of the matrix converter
and the inverter such as a dual active bridge converter.
However, the output voltage of the matrix converter isconverted from three-phase voltage. Therefore, the matrix
converter voltage is changed depend on grid frequency. It isnecessary only to consider from 0° to 60° of a three-phase gridvoltage.

Upper side
Lower side

Inverter

T

R

S

T
R

T
S1

S

T

S2

S3

S4

S1
S3

v1
v2
NVdc
0

-NVdc
-v2
-v1

(1)

where, Tsw is one switching period and L is an inductor
connected in series with the transformer, which is the value
converted to the primary side.

Transformer current itr

v1
Tsw

i1 = L Da 2v1 − NVdcT
D1 sw
i2 = i1 +L2v2 − NVdc
Tsw
D2
i3 = i2 +
L
2


Spr

T0 Ta T1 T2
i1
0

i2

Tb

T0

Ta T1 T2

Tb

T0

i3

i0
-i0
-i1

-i2

-i3

Tsw

Fig. 3. Fundamental voltage vector diagram to calculate the duty for thematrix converter.

Therefore, the relationship between the current command
I1, I2 and the duty ratio is expressed by (2) using the
instantaneous value of the transformer current.

Matrix converter voltage vmc
Inverter voltage Nvinv

v1
v2
NVdc
0

i1
i1 + i2
 I1 = 2 Da + 2 D1
 I = i2 + i3 D
2
 2
2

(2)

T0 Ta T1 T2

Tb

T0

i0

0

Ta T1 T2

Tb

T0

Tb

T0

i3

i1 i2

where, the combination of Da and D1 is not uniquelydetermined because there are no constraints. Therefore, thecoefficient a is defined as the ratio of Da and D1.
a = Da / D1

-NVdc
-v2
-v1

-i0
-i1

Transformer current itr

-i2

-i3

Tsw

(3)

(a) ZVS for inverter (i0 : Positive)

The coefficient a is adjusted according to the transition
power and the DC voltage. From (1)-(3), each duty ratio is
obtained as (4)-(7).

Matrix converter voltage vmc
Inverter voltage Nvinv

v1
v2
NVdc
0

4 I1 Lf sw
D1 =2
v1 (a + 2a + 1) − NVdc

(4)

-NVdc
-v2
-v1
T0

(

D2 = −b2 ± b2 2 − 4c2) D2(v1 (a + 1) − NVdc ) D1 + (v2 − NVdc ) D2
NVdc

T1 T2
i1

1

v (a + 1) − NVdc
I v (a + 1) 2 − NVdc
Q b2 = 1, c2 = 2 1
v2 − NVdc
I1
v2 − NVdc

Db =

Ta

(5)

0

i2

Tb
i3

T0

Ta

T1 T2

-i0
i0
-i1
Transformer current itr(6)

The switching pattern during T0 to Ta for the matrix
converter assumes that the commutation is from T-phase to Rphase. The drain-source voltage is not discharged by thecirculating current which is almost zero when the current pathin the matrix converter is changed from T-phase to R-phase.
The initial current is negative to discharge the drain-source
voltage in the matrix converter. However, the ZVS operationin the inverter does not achieve due to the negative initial
current. The discharge of the drain-source voltage of S1 forZVS does not achieved when S1 turns on after S2 turns off. In
order to achieve ZVS on the matrix converter side, the initial
current i0 needs to be negative. However, to achieve ZVS on
the inverter side, the initial current i0 must be set to positive.
In other words, depending on the current direction of i0, only
one of the converters achieves ZVS. Therefore, the duty cycle
for the grid current to sinusoidal is derived by considering i0.
In Chapter 2.A, the duty ratio is expressed based on the initial
current of zero. In next Chapter, the initial current is
considered to achieve the ZVS operation in the matrix
converter or inverter.
B. With Circulating Current
Fig. 4 (a) and (b) show the operation waveforms for the
initial current i0 in the positive and negative direction. In orderto superimpose the initial current i0 on each instantaneous
current, (2) is extended as (8).

-i2

-i3

Tsw
(b) ZVS for matrix converter (i0 : Negative)
Fig. 4. Medium-frequency voltage and current waveforms with initial
current to achieve ZVS.

Tsw
v1

i1 = L Da 2 + i0v1 − NVdc
T
D1 sw + i0
i2 = i1 +L2v2 − NVdc
T
D2 sw + i0
i3 = i2 +
L
2


(8)

The current-time product of each period also increases ordecreases depending on the initial current. Therefore, (3) is
extended to equation (9) when the initial current is taken into
account.
i0 + i1
i1 + i2
 I1 = 2 Da + 2 D1
 I = i2 + i3 D
2
 2
2

(9)

Substituting (8) and (3) into (9), each duty ratio can is
obtained as in (10)-(12).

D1 =−b1 + b12 − 4b1c1

2
4 Lf sw (a + 1)i0
I1
Q b1 =, c1 = −v1 (a + 1)2 − NVdc
(a + 1)i0

D2 =TABLE I.

(10)−b2 ± b2 2 − 4c2

2
2 D1 (v1 (a + 1) − NVdc ) + 4 Lf sw i0
Q b2 =
v2 − NVdc

Element

SymbolRated power

P

5 kW

Thre e-phase AC voltage

vac

200 V

Nominal DC voltage

Vdc

74 V

Grid f requency

f

50 Hz

Carr ier freque ncy

fsw

50 kHzL(%ZL)

20 µH(94%)

Grid side
LC filter
DC sideDead-time(12)
vmc 100[V/div]

Finally, the zero voltage period t0 is defined by (13).
D0 = 1 − ((a + 1) D1 + D2 + Db )Value

Leakage inductanceTurn r atio of transforme r(11)

4 I 2 Lf sw
Q c2 = −v2 − NVdc(v (a + 1) − NVdc ) D1 + (v2 − NVdc ) D2 + 4i0 Lf sw
Db = 1
NVdc

EXPERIMENTAL CONDITION.

N1:N2

3.3:1

Lac (%ZLac )190 µH(0.8%)

C ac (%YCac)
Ldc

15 µF(3.8%)
4.4 µHC dc

3 mF

Matrix converter

Tmc

Inve rterTinv

250 ns
400 nsvmc 100[V/div]

2[µs/div]

vinv 50[V/div]

2[µs/div]

vinv 50[V/div]

(13)
0

0

By substituting the sufficient current value for ZVS to theinitial current in (10)-(13), the duty ratios are obtained toachieve ZVS in the whole load range.
III. EXPERIMENTAL VERIFICATION
Table 1 shows the experimental conditions. The variationof the DC voltage is assumed to be plus or minus 20% of thenominal DC voltage. The nominal DC voltage is 74V because
it is the most convenient power supply voltage in ourlaboratory facilities. The nominal voltage can be set to 450 V
by changing the turn ratio from 3.3 to 0.5 when the DC voltage
range of 400 to 500 V is designed for EV applications. The
effectiveness of the proposed method is not affected by thechange in the DC voltage under the same boost ratio. The sumof the inductance which is connected in series with thetransformer is 20 µH. The switching device of the matrix
converter and inverter are SiC MOSFETs (C3M0030090K,
CREE) and Si MOSFETs (IXFH220N20X3, IXYS),
respectively. The turn-on and turn-off time of these switching
devices are different. Therefore, the dead-time of the matrix
converter and the inverter is also different. The switching
devices of the inverter are connected to parallel because thehigh RMS current occurs at the low DC voltage. The input
current distortion and the surge voltage of the matrix converter
side and the inverter side are evaluated.
A. Dead-time Compensation
Fig. 5 (a), (b) shows the operation waveforms without thedead-time compensation on the phase angle θ =0° and 20°.
The phase angle 0° is defined as the phase angle when the lineto-line voltage vuv is maximum. The duty ratio for the threephase isolated AC to DC converter using the matrix converter
is calculated by (4) to (6) with the initial current of zero.
However, the amplitude of the transformer current is clampedby 2 A during the zero voltage period T0 even though the initial
current of zero. The dead-time compensation is implemented
to eliminate the current error which is caused by the dead-timeerror.
Fig. 5(c), (d) shows the operation waveforms with thedead-time compensation. The dead-time error is compensateditr 10[A/div]0

itr 10[A/div]
0 ErrorError(a) θ = 0° without compensationvmc 100[V/div](b) θ = 20° without compensationvmc 100[V/div]

2[µs/div]

vinv 50[V/div]

0

0

0

2[µs/div]

vinv 50[V/div]

itr 10[A/div]
Zero-current(c) θ = 0° without compensation
Fig. 5.

itr 10[A/div]0
Zero-current(d) θ = 20° without compensation

Middile-frequency operation waveforms with dead-timeby adding or subtracting the dead-time Tmc of the matrix
converter to the duty ratio Ta because the amount of the deadtime compensation is constant regardless of the phase angle.
The amplitude of the transformer current becomes zero during
the zero voltage period to apply the dead-time compensation.
In the following sections, experimental results with the deadtime compensation are presented.
B. Surge Voltage Reduction by Circulating Current
Fig. 6 shows the drain-source voltage Vds, matrix converter
output voltage, inverter output voltage of the switching
devices S1 and S2 on the inverter side and Srp on the matrix
converter side at 1 kW (0.2p.u.). The transformer currents
during the zero-voltage period are clamped at each circulating
current with any current command of 0A, 3A, and -3A. When0

Vds of S rp 500[V/div]∆V=200V∆V=25VVds of S 2 100[V/div]0

Vds of S rp 500[V/div]

0

∆V=0V

Vds of S 2 100[V/div]0

Vds of S 1 100[V/div]

Vds of S 2 100[V/div]0

Vds of S 1 100[V/div]∆V=40VVds of S 1 100[V/div]

vmc 200[V/div]

vmc 200[V/div]

vmc 200[V/div]

vinv 100[V/div]

vinv 100[V/div]0
itr 10[A/div]00

itr 10[A/div]

itr 10[A/div]00
i0 =0Ai0 =3A
4[µs/div]i0 =-3A

4[µs/div](a) Hard switching of both converter (i0= 0A)

0

∆V=0V

vinv 100[V/div]0

Fig. 6.

Vds of S rp 500[V/div]0∆V=240V(b) ZVS of inverter (i0= 3A)

4[µs/div](c) ZVS of matrix converter (i0= -3A)

Comparison of surge voltage at matrix converter and inverter by changing initial current i0.

Upper side
Vgs[10V/div]

Upper side
Vgs[10V/div]0Upper side
Vds[50V/div]∆V:25V

Upper side
Vds[50V/div]

0

0

0

Upper side
Vgs[10V/div]
Upper side
Vds[50V/div]0

Lower side
Vgs[10V/div]

Lower side
Vds[50V/div]
Hard switching

ZVS

Zero-voltage

400[ns/div]0400[ns/div](a) Conventional method (i0= 0A)0

Lower side
Vgs[10V/div]

Lower side
Vgs[10V/div]
Lower side
Vds[50V/div]

Lower side
Vds[50V/div]

Lower side
Vds[50V/div]

0

0

0

Upper side
Vds[50V/div]

∆V:5V

0

0

ZVS

0 Lower side
V gs[10V/div]

Upper side
Vgs[10V/div]0

Zero-voltage

0400[ns/div]

ZVS

400[ns/div](b)Proposed method (i0= 3A)

Fig. 7. Zero voltage switching by circulation current i0.

the conventional circulation current is 0 A, the surge voltage
generated at the drain-source voltage of S2 on the inverter side
is 25 V, which is about 1.3 times higher than the rated DC
voltage. The surge voltage of the switching devices on thematrix converter side is 200V. These are caused by the hardswitching due to the lack of the instantaneous current todischarge the drain source-voltage. The recovery current
flows and the surge voltage is generated due to this hardswitching.
Fig. 6 (b) shows the operation waveforms with thepositive direction of the circulating current (3 A). The surge
voltage of S2 is suppressed by achieving ZVS when the drainsource voltage of the switching device is completely
discharged by the circulating current in the inverter side. On
the other hand, the surge voltage in the matrix converter sideincreases from 200 V to 240 V because the instantaneous
current at turn-off is increased by the circulating current.
Fig. 6 (c) shows the surge voltage with the negative
direction of the circulating current (-3 A). The surge voltage
generated at Srp is suppressed to achieve ZVS at the matrix
converter side. On the other hand, the surge voltage generated
on the inverter side increases from 25V to 40V. Next section,
the surge voltage and efficiency are evaluated when ZVS isachieved on the inverter side with the initial current of 3A.
Fig. 7 shows the gate-source and the drain-source voltage
waveforms of S1 and S2 when these switches turn-on and turnoff, respectively. The transformer current is equal to thecirculating current when the transition power reference is zero.

The surge voltage occurs at the switching device S1 in theupper side under zero initial current. On the other hand, thesurge voltage of S1 is suppressed by the circulating current of
3 A even though no-load state.
C. Converter Characteristics
Fig. 8 shows the relationship between the transmission
power and the surge voltage generated at S1 normalized by
the rated DC voltage. In the DAB matrix converter, the drainsource voltage is completely discharged by the circulating
current in the heavy load condition. However, theinstantaneous current is not enough to the ZVS condition in
the light load condition. In the proposed method, thecirculating current is not changed by the transition power andthe AC and DC voltage amplitudes. Therefore, ZVS isachieved in the whole load condition by setting thecirculating current to 3A for ZVS of the inverter side. Figure5(b) shows the efficiency characteristics when the circulating
current is set to 0 A and 3 A. The proposed method achieves
ZVS of the inverter side. However, the RMS value of thetransformer current increases to increase the circulating
current by applying the proposed method. The efficiency
characteristic in the light-load is equal to or higher than thatof the conventional method.
Fig. 9 shows the efficiency characteristics under the rated
DC voltage. The co-efficient a which is decided by (3) ischanging based on the transition power. Here, the maximum
transmission power is determined by the coefficient a.
Therefore, it must be changed appropriately according to theFig. 9(b) and 9(c) show the transformer current waveforms
for a = 0.2 and 2, respectively. The transmission power is thesame at 0.2 p.u. (1 kW). The duty ratio Da increases with thecoefficient of 2 compared to 0.2. Therefore, the increase in thecurrent peak value causes the conduction loss to increase. By
selecting the optimum ratio according to the transmission
power, the envelope of the largest part of the efficiency
characteristics shown in Fig. 9(a) is assumed to be theefficiency characteristics of the circuit under consideration.

The maximum transmission power varied with the DC
voltage could be transmitted at the 0.9 p.u. of rated power.
The system efficiency at the low voltage is lower than one atnominal voltage because the RMS current at the secondaryside increases. The current distortion was suppressed to lessthan 5.0% at the rated power of 0.4 p.u. or more, regardless of
the DC voltage.
The suppression of the grid current distortion and ZVS areachieved simultaneously in whole load regions. Therefore, theusefulness of the proposed surge voltage reduction method isdemonstrated. In addition, the isolated three-phase matrix ACDC converter using the matrix converter with the proposed
method is operated under a wide DC voltage range which isachieved by the buck-boost and boost operation.

[2]

[3]

V. Vlatkovic, D. Borojevic, X. Zhuang and F. C. Lee, "Analysis anddesign of a zero-voltage switched, three-phase PWM rectifier withpower factor correction," PESC '92 Record. 23rd Annual IEEE PowerElectronics Specialists Conference, Toledo, Spain, 1992, pp. 13521360 vol.2.
V. Vlatkovic, D. Borojevic and F. C. Lee, "A zero-voltage switched,
three-phase isolated PWM buck rectifier," in IEEE Transactions on
Power Electronics, vol. 10, no. 2, pp. 148-157, March 1995.
Z. Yan, M. Jia, C. Zhang and W. Wu, "An Integration SPWM Strategy
for High-Frequency Link Matrix Converter With Adaptive
Commutation in One Step Based on De-Re-Coupling Idea," in IEEE
Transactions on Industrial Electronics, vol. 59, no. 1, pp. 116-128, Jan.
2012.

Proposed method :i0 = 3[A]
0.5
0 0

0.05

a= 0.20.20
0.25
1p.u. = 5.0 kW

0.10
0.15
Power[p.u.]

Proposed method :i0 = 3[A]Efficiency [%]96
94

Conventional method : i0 = 0[A]9290a= 0.2
88
0.0

0.1

0.2

0.3

0.4

1p.u. = 5.0 kW

Power [p.u.]

0.5(b) Efficiency characteristics
Fig. 8. Evaluation results of characteristics with each initial current.

98
Fig. 9(b)

a=0.1

a=0.2

96

a=0.5

94

a=1
a=2

92
Fig. 9(c)90

In this paper, the ZVS method using the circulating
current is proposed to reduce the surge voltage in either thematrix converter side or the inverter side. ZVS of the inverter
side or the matrix converter side is selected by the directionof the circulating current. Therefore, the surge voltage is
suppressed to less than 1.1 times of the rated DC voltage.
Furthermore, the grid current THD is suppressed to less than
3.0% from the duty ratio calculation considering the increaseof the circulating current for ZVS.

[1]

1.0

98

IV. CONCLUSIONSREFERENCES

Conventional method : i0 = 0[A]1.5(a) Surge voltage characteristics

Efficiency [%]Fig. 10(a) and 10 (b) show the efficiency and THD
characteristics under the DC voltage up to 20% at the rated
power 5-kW. Assume that the battery is connected to the load,
the efficiency and THD characteristics of the isolated threephase matrix DAB AC-DC converter are evaluated when the
DC voltage changes by plus or minus 20% against the nominal
DC voltage. The DC voltage is generated by a DC power
supply (PSB 9000 3U, EA - Elektro Automatik GmbH & Co.
KG). It is evaluated up to the maximum transmission power at
each voltage condition.

2.0

Surge voltage nomilized bynominal DC voltage[p.u.]load. The optimization for coefficient a does not discuss toimprove the converter efficiency in this paper. In this paper,
the efficiency characteristics were obtained by varying thecoefficient a from 0 to 2. The maximum efficiency of 96.0%
was achieved when a = 0.2.

88

0

0.2

0.4

0.6

0.8

Power [p.u.]1

1p.u. = 5.0 kW(a) Efficiency characteristics with nominal DC voltage.
vmc 250[V/div]
vinv 100[V/div]

vmc 250[V/div]
vinv 100[V/div]
0

0

0

0
itr 10[A/div]

itr 10[A/div]
4[µs/div]

4[µs/div](b) Da/D1 = 0.2(c) Da/D1 = 2

Fig. 9. Relationship between ratio Da /D1 and efficiency characteristics.

[4]

98

Efficiency [%]96
Vdc = 1.2Vnom
Vdc = 1.0Vnom

94
92

Vdc = 0.8Vnom

90
88

a=1
Vnom = 74V

0

0.2

0.4

0.6

0.8

Power [p.u.](a) Efficiency characteristics

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9
8
7
6
5
4
3
2
1
0

1

1p.u. = 5.0 kW

Vdc = 0.8Vnom
Vdc = 1.0Vnom

Vdc = 1.2Vnom
a=1
Vnom = 74V

0

0.2

0.4

0.6

Power [p.u.](b) THD characteristics

0.8

1

1p.u. = 5.0 kW

Fig. 10. Efficiency and THD characteristics with DC voltage variation ofplus or minus 20%.

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