UNIVERSIDAD DE OVIEDOTESIS DOCTORAL
16/04/2004

CERN-THESIS-2004-027

Departamento de Ingeniería Eléctrica, Electrónica, de Computadoresy Sistemas

EMC: A mapping for the CMS
experiment

Fernando José González Arteche
2004Acknowledgements

In a few lines I would like to thank to everybody that has helped me in thedevelopment of this thesis. During the last three years that I have been working on
my PhD I have received the support of a lot of people that I would like to thanknow. I hope I do not forget any of them along the following paragraphs.
First of all I would like to thank to my supervisors Dr. Peter Sharp and Prof. Javier
Gómez-Aleixandre Fernández for their enthusiastic support during the developmentof the thesis. Thanks Peter for your help, you have always found time to talk to me;
your office's door has always been open for me, no matter time or day. Thanks Javier
for your help from the Electrical and Electronic Engineering Department of the
University of Oviedo. Thanks to answer all my questions as well as for dealing withall official issues that I had with the University.
From FERMILAB, I would like to thank Dr. Claudio Rivetta. I do not know how tothank you. I think there are not enough words to thank you for the hundreds ofe-mails that you have written to me in these years, for the long VRVS-conferences todiscuss technical aspects of the project, for supporting me during the bad moments,
for sharing with me your engineering knowledge, for your insights and support
during the writing of the thesis, for your willingness to proof-read the chapters andfor your fruitful suggestions and technical discussions. "Muchas gracias flaco, te estaré
eternamente agradecido".
From CERN, I would like to express a warm "thank you" to Dr. Fritz Szoncso for his
unconditional support during the bad moments, enthusiasm and fruitful technicaldiscussions during coffee breaks. Also, I would like to thank you, David Nisbet foryour fruitful discussions about technical aspects and thoughtful comments on
different versions of the thesis. I would like to thank you for being patient andreading carefully the thesis, giving to the document the final English configuration.
I would like to thank all members of the electronic pool too. Especially, I would liketo thank Bruno Allongue and Philippe Fontaine-Vive-Roux for their support during the
EMC tests carried out at CERN. Great thanks as well to Dr. Horst Breuker, Dr. RobertHammarstrom and Virginia Prieto for the help that I have received from them during
the thesis.
From the EMC department of the European Space Agency, I would like to thank Dr.
Antonio Ciccolella and Philippe Langet for the three-day meeting they spent with me.
Especially, I thank you, Antonio, for devoting part of your time to talk to me bytelephone or write e-mails, where you kindly answered me all technical questions.

Acknowledgements

You and your group have always been ready to offer me support and to give me
useful suggestions.
Great thanks to the HCAL US group, Ms. Terry Shaw, Mr. Sergey Los, Dr. John Elias,
Terry Grozis and to all other members for their support during the EMC tests
performed at FERMILAB. Also from HCAL group I would like to thank to
Dragoslav-Laza Lazic for his help during the preparation and coordination of the trip
to FERMILAB.
From University of Oviedo I would like to thank Prof. Guillermo Ojea Merín and Dr.
Alberto Diez González for their support from the Electrical and Electronics
Engineering Department of the University of Oviedo. Thanks a lot Alberto for the"financial" help that I have received from your group.
I would like to thank my dear friends Eduardo (gracias por haber tenido la paciencia deleer la tesis), Silvia and Barbara for their support during the whole development of
the Thesis. Also, great thanks family Guerrero-van der Ploeg for your support.
To my family, gracias mama, Nacho, Yoyi, Natalia, Jofre y Atena por vuestra ayuda
incondicional todos estos años que he pasado lejos de casa.
Finally, I would like to thank a very special person. Ana, thanks for yourunconditional support, love, and basically just for being yourself. Without yoursupport I think that I would not have finished this project.

Contents
Introducción - Versión en castellano .

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. 1

Introduction .

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Chapter 1
CMS detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1.1 The Large Hadron Collider (LHC) . . . . . . . . . . . . . . .
1.2 The Compact Muon Solenoid (CMS) experiment . . . . . . . . .
1.3 Tracker Sub-system . . . . . . . . . . . . . . . . . . . .
1.3.1 Pixel detector . . . . . . . . . . . . . . . . . . .
1.3.2 Silicon detector . . . . . . . . . . . . . . . . . .
1.3.3 Tracker Readout Electronics . . . . . . . . . . . . . .
1.3.3.1 APV Front End circuit. . . . . . . . . . . . .
1.3.3.2 Optical Link . . . . . . . . . . . . . . . .
1.3.3.3 Front End Driver . . . . . . . . . . . . . .
1.3.3.4 Control System . . . . . . . . . . . . . . .
1.3.3.5 Low and High voltage PS distribution system. . . .
1.4 Electromagnetic Calorimeter (ECAL) . . . . . . . . . . . . . .
1.4.1 ECAL Read out electronics . . . . . . . . . . . . . .
1.4.2 Pre-shower detector . . . . . . . . . . . . . . . . .
1.4.2.1 Pre-shower read-out electronics . . . . . . . .
1.5 MUON chambers sub-system . . . . . . . . . . . . . . . . .
1.5.1 Muon Sub-system read-out electronics . . . . . . . . . .
1.6 Super-conducting Magnet Sub-system . . . . . . . . . . . . .
1.6.1 Super-conducting coil . . . . . . . . . . . . . . . .
1.6.2 Power supply system of Super-conducting solenoid . . . . .
1.7 HCAL sub-system . . . . . . . . . . . . . . . . . . . . .
1.7.1 System generalities . . . . . . . . . . . . . . . . .
1.7.2 HCAL FEE . . . . . . . . . . . . . . . . . . . .
1.7.2.1 Introduction . . . . . . . . . . . . . . . .
1.7.2.2 System configuration . . . . . . . . . . . .
1.7.2.3 Front-end electronics . . . . . . . . . . . .
1.7.2.4 Analogue signal conditioning . . . . . . . . .
1.7.2.5 Dynamic range . . . . . . . . . . . . . . .
1.7.3 FEE Components . . . . . . . . . . . . . . . . . .
1.7.3.1 Overview . . . . . . . . . . . . . . . . .
1.7.3.2 Analogue conditioning and digitising ASIC-QIE . .
1.7.3.3 Channel control ASCI (CCA) . . . . . . . . .
1.7.3.4 Data links . . . . . . . . . . . . . . . . .
1.7.3.5 Linear Rad-Tolerant Voltage Regulators . . . . .
1.7.4 Front-end Electronics Readout Boxes (RBX). . . . . . . . .
1.7.4.1 Channel distribution per RBX . . . . . . . . .
1.7.4.2 RBX layout . . . . . . . . . . . . . . . .
1.7.4.3 Power requirements per RBX . . . . . . . . .
1.8 PS distribution system of CMS detector . . . . . . . . . . . . .
1.8.1 Introduction . . . . . . . . . . . . . . . . . . . .

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1.9 Summary .
1.10 References .

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. 44

Chapter 2
Grounding & Shielding . . . . . . . . . . . . . . . . . . . . . . . . . .
2.1 Grounding generalities . . . . . . . . . . . . . . . . . . .
2.2 Grounding for fault and personnel protection. . . . . . . . . . .
2.2.1 System grounding or earthing . . . . . . . . . . . . .
2.2.2 Equipment grounding . . . . . . . . . . . . . . . .
2.2.2.1 Standard equipment grounding . . . . . . . .
2.2.2.2 Isolated grounding . . . . . . . . . . . . .
2.3 High frequency grounding configuration . . . . . . . . . . . .
2.3.1 Multipoint and single point grounding connections . . . . .
2.3.1.1 Single point . . . . . . . . . . . . . . . .
2.3.1.2 Multipoint . . . . . . . . . . . . . . . .
2.3.2 Bonding and straps . . . . . . . . . . . . . . . . .
2.3.3 Signal reference structure for non-continuous area . . . . .
2.4 Shielding . . . . . . . . . . . . . . . . . . . . . . . .
2.4.1 Introduction . . . . . . . . . . . . . . . . . . . .
2.4.2 Electromagnetic field shielding . . . . . . . . . . . . .
2.4.3 Electric field shielding . . . . . . . . . . . . . . . .
2.4.4 Magnetic field shielding . . . . . . . . . . . . . . .
2.4.5 Cable shielding . . . . . . . . . . . . . . . . . .
2.5 CMS grounding . . . . . . . . . . . . . . . . . . . . . .
2.6 Summary . . . . . . . . . . . . . . . . . . . . . . . .
2.7 References . . . . . . . . . . . . . . . . . . . . . . . .
Chapter 3
Conducted noise . . . . . . . . . . . . . . . . . . . . . . .
3.1 RF conducted noise . . . . . . . . . . . . . . .
3.2 Switching mode power supplies . . . . . . . . . . .
3.2.1 Conducted RF emission of SMPS . . . . . . .
3.2.2 Conducted noise measurements . . . . . . .
3.2.2.1 Experimental set-up . . . . . . . .
3.2.2.2 Measurement results . . . . . . .
3.3 Filter selection . . . . . . . . . . . . . . . . .
3.3.1 Non ideal behavior of components. . . . . . .
3.3.1.1 Resistor . . . . . . . . . . . .
3.3.1.2 Capacitors . . . . . . . . . . .
3.3.1.3 Inductors . . . . . . . . . . . .
3.3.1.4 Common mode chokes . . . . . . .
3.3.2 Insertion loss . . . . . . . . . . . . . .
3.3.3 Filter characterization . . . . . . . . . . .
3.3.4 Filter performance . . . . . . . . . . . .
3.3.4.1 Output filter performance . . . . . .
3.3.4.2 Input filter performance . . . . . .
3.4 Stability problems with SMPS - Negative input impedance .
3.5 Harmonics . . . . . . . . . . . . . . . . . . .
3.6 Summary . . . . . . . . . . . . . . . . . . .
3.7 References . . . . . . . . . . . . . . . . . . .

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Chapter 4
Noise Propagation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125
4.1 Introduction to Multi-conductor Transmission lines (MTL) . . . . . . . . . . 125
4.2 Definitions of per-unit length parameters . . . . . . . . . . . . . . . . 127
4.2.1 Per unit length inductance matrix . . . . . . . . . . . . . . . . 127
4.2.2 Per unit length capacitance matrix . . . . . . . . . . . . . . . 129
4.2.3 Per unit length resistance matrix . . . . . . . . . . . . . . . . 131
4.2.4 Per unit length conductance matrix . . . . . . . . . . . . . . . 131
4.3 Parameters measurement procedure . . . . . . . . . . . . . . . . . . 132
4.3.1 Inductance . . . . . . . . . . . . . . . . . . . . . . . . 132
4.3.2 Capacitance . . . . . . . . . . . . . . . . . . . . . . . . 135
4.3.3 Resistance . . . . . . . . . . . . . . . . . . . . . . . . 138
4.4 Frequency domain solution of MTL systems. . . . . . . . . . . . . . . . 140
4.4.1 Solution of MTL incorporating the terminal conditions. . . . . . . . 144
4.5 Noise propagation in cables . . . . . . . . . . . . . . . . . . . . . 146
4.5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . 146
4.5.2 Common mode noise effects . . . . . . . . . . . . . . . . . . 149
4.5.3 Differential mode noise effects . . . . . . . . . . . . . . . . . 152
4.5.4 Summary of noise propagation . . . . . . . . . . . . . . . . . 155
4.5.5 CM and DM mode noise propagation in HCAL
power supply distribution system . . . . . . . . . . . . . . . 156
4.5.5.1 Introduction . . . . . . . . . . . . . . . . . . . . 156
4.5.5.2 Common mode noise propagation . . . . . . . . . . . 157
4.5.6 Differential mode noise propagation . . . . . . . . . . . . . . . 160
4.6 Shield models . . . . . . . . . . . . . . . . . . . . . . . . . . 162
4.6.1 Surface transfer impedance and admittance . . . . . . . . . . . . 162
4.6.2 Solution for the cable including the shield in the frequency domain . . . 164
4.6.3 Susceptibility of HCAL power cable to shield currents . . . . . . . . 167
4.6.3.1 Introduction . . . . . . . . . . . . . . . . . . . . 167
4.6.3.2 Common mode . . . . . . . . . . . . . . . . . . 169
4.6.3.3 Differential mode . . . . . . . . . . . . . . . . . . 170
4.7 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . 172
4.8 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . 172
Chapter 5
Susceptibility at early design stage . . . . . . . . . . . . . . .
5.1 Introduction . . . . . . . . . . . . . . . . . .
5.2 Considerations about the noise in the FEE . . . . . . .
5.3 Thermal noise . . . . . . . . . . . . . . . . .
5.4 Susceptibility to CM currents . . . . . . . . . . . .
5.5 External electromagnetic fields . . . . . . . . . . .
5.5.1 Introduction . . . . . . . . . . . . . . .
5.5.2 Uniform plane-wave excitation of the line. . . .
5.5.3 Susceptibility to Electromagnetic fields of HF FEE.
5.6 Summary . . . . . . . . . . . . . . . . . . .
5.7 References . . . . . . . . . . . . . . . . . . .
Chapter 6
RF Susceptibility Test of FEE . . . . . . . . .
6.1 Introduction . . . . . . . . . . .
6.1.1 Injection of shield currents . .

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6.2
6.3

6.4

6.5
6.6

6.7
6.8

6.1.2 Injection of CM currents in the central conductors. . .
6.1.3 Injection of DM currents in the central conductors . .
6.1.4 Injection of CM+DM currents in the central conductors.
Experimental set-up . . . . . . . . . . . . . . . . .
6.2.1 HCAL FEE prototype . . . . . . . . . . . . .
Susceptibility to shield currents . . . . . . . . . . . . .
6.3.1 Noise distribution . . . . . . . . . . . . . .
6.3.2 Transfer function . . . . . . . . . . . . . . .
6.3.3 Grounding configurations. . . . . . . . . . . .
Susceptibility to CM power currents . . . . . . . . . . .
6.4.1 Noise distribution . . . . . . . . . . . . . .
6.4.2 Transfer function . . . . . . . . . . . . . . .
6.4.3 Common mode filter effects. . . . . . . . . . . .
Susceptibility to DM power currents . . . . . . . . . . .
DM and CM noise contributions to FEE output . . . . . . .
6.6.1 Noise distribution and TF of CM + DM currents. . . .
6.6.2 External noise contributions to the overall FEE noise. .
6.6.3 Normalized noise level required at the output of the PS
Summary . . . . . . . . . . . . . . . . . . . . .
References . . . . . . . . . . . . . . . . . . . . .

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Chapter 7
Susceptibility to transient perturbation . . . . . . . . . . . . . . . . . . . . . . . . 237
7.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . 237
7.2 Voltage dips and short interruptions . . . . . . . . . . . . . . . . . . 240
7.2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . 240
7.2.2 HCAL FEE susceptibility test to voltage dips and short interruptions. . . 241
7.3 Surge or slow transients . . . . . . . . . . . . . . . . . . . . . . . 244
7.3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . 244
7.3.2 Transients in DC distribution . . . . . . . . . . . . . . . . . 246
7.3.2.1 The Tracker DC power distribution system . . . . . . . . 247
7.3.2.2 Multi-module DC distribution . . . . . . . . . . . . . 256
7.3.3 Surge test for the CMS detector. . . . . . . . . . . . . . . . . 264
7.3.3.1 Introduction . . . . . . . . . . . . . . . . . . . . 264
7.3.3.2 Characterization of the HCAL FEE to surge test. . . . . . . 266
7.3.3.3 Protection stages for the CMS detector. . . . . . . . . . . 270
7.4 Burst or fast transients . . . . . . . . . . . . . . . . . . . . . . . 271
7.5 Quenching effects . . . . . . . . . . . . . . . . . . . . . . . . . 273
7.6 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . 275
7.7 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . 277
Conclusions

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Thesis contribution and future work

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Conclusiones - Versión en castellano

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Objetivos de la tesis & Futuros Trabajos -Versión en castellano .
Appendix A
MTL . . . . . . . . . . .
A.1 Internal impedances - Skin effect

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A.2 Characteristic impedance . . . . . . . . . . . .
A.2.1 CM and DM characteristic impedance matrix .
A.3 MTL programs validation . . . . . . . . . . . .
A.3.1 Ribbon cable . . . . . . . . . . . . .
A.3.2 Field incident . . . . . . . . . . . . .
A.4 References . . . . . . . . . . . . . . . . . .
Appendix B
Transients. . . . . . . . . . . .
B.1 Dynamic load impedances of Tracker FEE .
B.2 Transient lumped model validation . . .
B.3 ATO fuse data sheet . . . . . . . . .
B.4 TVS validation . . . . . . . . . . .
Appendix C
Current probes.

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Introducción - Versión en castellano

El Laboratorio Europeo de Física de Partículas, CERN, es el mayor centro de física
de partículas del mundo. Dicho centro, formado por veinte países europeos, está
situado en la frontera franco-suiza, al oeste de Ginebra. En el CERN se realiza
investigación básica, con el objetivo de descubrir de qué está hecho nuestro
Universo y a qué leyes obedece. Para ello, se estudia el comportamiento de loscomponentes más pequeños de la naturaleza: las partículas elementales. Con estefin, se realizan diferentes tipos de experimentos científicos utilizando aceleradores ydetectores de partículas. En los aceleradores se llevan las partículas hasta altos
niveles de energía y después se las hace colisionar entre ellas, pudiendo observarselo que ocurre en los detectores.
Actualmente en el CERN se está realizando la construcción del más potente y largoacelerador de partículas del mundo llamado Large Hadron Collider (LHC). Este
acelerador tiene como objetivo profundizar en el estudio de la estructura de lamateria, recreando las condiciones que existieron en el Universo instantes despuésdel "Big Bang". Junto con el acelerador, se está procediendo a la construcción de 4
detectores de partículas. El detector llamado CMS (Compact Muon Solenoid) es
uno de ellos. Las dimensiones de este detector son de unos 30 metros de largo por
25 de diámetro, con un peso total aproximado de 13000 toneladas.
El detector CMS esta dividido en 5 subsistemas, los cuales se encuentran
emplazados en diferentes niveles dentro de la estructura del detector. Cada
subsistema tiene una función diferente y maneja diferentes niveles de señal ypotencia, siendo alimentados independientemente. En general, los experimentos defísica de altas energías tienen como función identificar partículas y medir su energía
usando diversos tipos de sensores y dispositivos ópticos. En el experimento de
CMS, la señal eléctrica detectada es amplificada y procesada por el FEE (front end
electronics), el cual se comunica con el sistema de adquisición de datos mediante
fibra óptica. Dicho sistema se encuentra localizado a unos 120 metros del detector.
La electrónica del detector CMS esta diseñada para procesar señales de pocos
milivoltios para su posterior digitalización sincronizada a 40 MHz. Parte de esta
electrónica se encuentra situada en la sección interna del detector teniendo que
soportar una elevada radiación protónica y un altísimo campo magnético de unos 4
Tesla.
Todos los datos del FEE son transmitidos al sistema de adquisición por fibra óptica,
con lo que se evitan problemas de integridad de señal generados por ruido. El
problema aparece intrínsecamente en el detector debido al elevado numero de1

Introducción - Versión en castellanointeracciones electromagnéticas entre los diferentes subsistemas. Los principales
problemas de interferencia electromagnética se pueden encontrar en la conexión
entre el sistema de alimentación y el FEE y en el propio FEE. Los elementos más
importantes a tener en cuenta antes de integrar la electrónica del detector son, por
un lado, la definición de los niveles de ruido y transitorios dentro de cadasubsistema, y por otro, cómo imponer estos niveles al sistema electrónico mediante
medidas normalizadas. CMS necesita generar una especificación tanto para laemisión como para la inmunidad al ruido que permitan integrar el detector de una
forma segura. Describir el escenario electromagnético de CMS es de crucial
importancia para fijar y resolver problemas referentes
a interferencias
electromagnéticas, puesta a tierra y blindajes. Esta tesis se centra en los estudios de
EMC realizados en el experimento CMS con el fin de definir la base de una futura
norma de EMC para el detector. Fundamentalmente estos estudios están enfocadosa la cuantificación de la sensibilidad del FEE al ruido conducido acoplado a travésde los cables de entrada y salida de la electrónica, así como del ruido generado porlas fuentes de alimentación. Asimismo, se consideran la magnitud y los efectos delos transitorios en el sistema de potencia.
Los estudios de EMC en CMS se han basado en los estándares industriales,comerciales y militares existentes, así como en reglas de las agencias espaciales(ESA y NASA). Dichos estándares han sido adaptados a los requerimientos de laelectrónica de CMS basándose en medidas y estudios de EMC en prototipos de loscomponentes del sistema de distribución. El principal objetivo de estos estudios es
definir buenas topologías en los componentes del sistema, niveles de emisión einmunidad, así como niveles de transitorios. Estos análisis permitirán realizar laintegración del detector de forma que el correcto funcionamiento del detector esté
asegurado. Además, esta tesis es la base de futuros estudios que posteriormentedarán lugar a los futuros estándares de EMC en física de altas energías.
La gran complejidad del detector CMS hace que no se pueda analizar cada sistema
en detalle, sin embargo, las reglas generales obtenidas a partir algunos subsistemas,
como son el Tracker y HCAL, pueden ser extendidas al resto del experimento. Losestudios de EMC se han centrado en el sistema de distribución de potencia, alconsiderar CMS como un sistema aislado en donde la única conexión galvánica es a
través de los cables del sistema de potencia. Para ello, se ha estudiado de forma
simplificada, habiéndose desglosado en tres partes: FEE, cables y fuentes dealimentación.
El FEE es el componente más sensible del sistema y será caracterizado por lasusceptibilidad al ruido conducido. En las fuentes de alimentación es necesario
caracterizar sus niveles de emisión. Esto es especialmente importante en losconvertidores DC-DC, los cuales, debido a sus altas emisiones radiadas y
conducidas, necesitan de un adecuado filtrado y blindaje con el fin de reducir sus
emisiones a niveles que puedan ser tolerados por la electrónica. Los cables son
generalmente el medio de acoplamiento del ruido entre unos sistemas y otros.

2Introducción - Versión en castellano

Afortunadamente, la tecnología actual permite reducir el volumen de cables en
CMS, al estar conectados el FEE y el sistema de adquisición de datos por fibraóptica. Esto reduce notablemente el nivel de inmunidad del FEE, lo que permite una
simplificación notable de la integración de la electrónica del detector. No obstante,el estudio de los efectos del ruido de modo común y diferencial en cables largos depotencia constituye un importante punto a analizar. Estos efectos son cruciales parala transmisión de ruido dentro del FEE y han de ser tenidos en cuenta en el diseñodel cableado y en la selección del tipo de cable (cable trenzado, blindado) a instalar
en el detector.
La tesis doctoral ha sido desarrollada en tres etapas diferentes, las cuales
corresponden, aproximadamente, a los trabajos realizados en cada uno de los tresaños que ha durado el trabajo realizado en el CERN. Durante el primer año, la tesis
se ha centrado en los estudios del sistema de distribución de potencia de CMS, enconcreto en los subsistema Tracker y HCAL, y en el diseño, de la electrónica deestos sistemas. Se realizaron una serie de ensayos preliminares para evaluar lasemisiones conducidas de los convertidores DC-DC. Los resultados obtenidos han
sido usados para determinar la atenuación de ruido necesaria en el sistema.
En el caso del sub-detector Tracker, se piensan instalar las fuentes de alimentación
en la sala anexa a la caverna (counting room). Esto hace que la fuentes esténsituadas a unos 100 o 120 metros de distancia y la energía sea suministrada a travésde unos cables especiales. Los principales parámetros y características del cable hansido definidos basándose en medidas y en modelos de cables. El principal objetivode este análisis ha sido definir los modelos adecuados que posteriormente nos
permitan estudiar la estabilidad del sistema de distribución y la respuesta a
perturbaciones en la corriente de carga, así como la propagación del ruido de alta
frecuencia entre las fuentes de alimentación y el FEE.
Durante el segundo año, los estudios se han enfocado en el desarrollo e
implementación de los ensayos de EMC, que constituirán la base de la propuestadel plan de EMC para CMS. Los ensayos han sido diseñados basándose en
recomendaciones de los ingenieros de la Agencia Espacial Europea (ESA). Estos
ensayos fueron llevados a cabo en el laboratorio FERMILAB en Chicago (USA).
El FEE del subsistema "Very Forwad HCAL" de CMS presenta una conexión
bastante especial entre el detector y el FEE. Esta conexión se ha usado para estudiar
la inmunidad de la electrónica a las señales de ruido de modo común, así como la
inmunidad a campos electromagnéticos externos, durante la fase de diseño. El
efecto de "crosstalk"en cables multi-conductores y el acoplamiento de campos
externos y tensiones de modo común son igualmente analizados en este punto.
CMS planea distribuir potencia hasta la periferia del detector mediate un sistema
trifásico de 400 Hz. En la periferia, convertidores AC-DC transformarán la tensión
alterna en la apropiada baja tensión continua necesaria para operar la electrónica

3

Introducción - Versión en castellanodel detector. También fueron analizadas las implicaciones que tiene este tipo dedistribución en CMS.
Finalmente, los resultados obtenidos mediante los ensayos de EMC realizados en el
FERMILAB han sido analizados durante el último año de tesis. La susceptibilidad
del FEE a perturbaciones electromagnéticas ha sido definida. También durante este
año se han analizado los efectos de la propagación de ruido en alta frecuencia a
través de cables de potencia y sus implicaciones en la especificación final de losniveles de emisión de las fuentes de alimentación. Con el fin de completar la
caracterización de las emisiones de las fuentes de alimentación, así como de losfiltros a incluir en ella, se realizaron unos ensayos adicionales.
El documento de tesis esta organizado de la siguiente forma:• Descripción del detector CMS (Capítulo 1)• Puesta a tierra del detector. (Capítulo 2)• Emisiones conducidas y filtrado de convertidores DC-DC. (Capítulo 3)• Efectos en la propagación del ruido en cables de potencia. (Capítulo 4)• Inmunidad del ruido en la fase de diseño. (Capítulo 5)• Ensayo de susceptibilidad del FEE. (Capítulo 6)• Transitorios en el sistema de distribución de CMS. (Capítulo 7)4

Introduction

The Compact Muon Solenoid (CMS) is one of the four high-energy physics
experiments under construction at CERN for the Large Hadron Collider (LHC)
accelerator. The dimensions of the CMS detector are about 30 meters long and25-meter diameter, with a total weight of around 13000 tons.
The CMS detector is divided in five sub-systems located at different layers of thestructure. Each sub-system has distinct objectives and processes different signal andenergy levels. In general, high-energy physics (HEP) experiments identify particle
interactions and measure their energy using sensitive detection devices such as
silicon and pixel detectors, wire chambers, scintillators and optical devices. In the
CMS experiment, the detected electrical signal is amplified and processed by thefront-end electronics (FEE), which communicate, via optical links, with theacquisition system located 120 meters away from the detector. The read-out
electronics of the CMS experiment are designed to process signals of a few mV
digitizing them synchronously at 40 MHz. Part of this electronics is located inside ofa harsh environment with particle radiation and a DC magnetic field of 4 Tesla.
All the data from the FEE is transmitted to the acquisition system via optical fibers,
avoiding that the electromagnetic interference (EMI) is coupled to the FEE fromoutside of the detector. The problem may appear in the detector itself, where a large
amount of electromagnetic interactions exist among the electronic sub-systems.
These interactions can be mainly found in the connection between the FEE and thepower supply distribution and in the FEE itself. The two most important issues toaddress before integrating the electronic system into the detector are first, thedefinition of the maximum levels of the radio-frequency noise and the transient
signals such that the FEE in each sub-system can operate achieving the performancegoals and second, the definition of how to impose those levels to the electronicsystem using normalized measurements. Within this framework, CMS needs togenerate a specification of both emission and immunity noise limits for theelectronic equipment to be integrated into the experiment. Normalized tests for theequipment have to be defined to comply with those limits. Describing theelectromagnetic (EM) environment of CMS is of major importance for determining
and solving problems related to electromagnetic interference, grounding andshielding. This thesis addresses the EMC studies conducted for the CMS
experiment to define the baseline of the future CMS EMC policy. The EMC studies
are mainly focused on the quantification of the FEE sensitivity to conductive noise
coupled through the input/output cables and the noise generated by power5Introduction

supplies. Also, the magnitude and effects of transient signals disturbing the power
supply system are introduced.
The EMC studies in CMS are based on existing industrial, commercial and military
standards and rules applied by aerospace agencies. These standards have been
adapted to the CMS electronic systems based on EMC measurements and EMC
studies, which include immunity test conducted on the FEE and emission tests
performed in different components of a power supply distribution systemprototype. The goal of these EMC studies is to define good topologies, emission andimmunity levels and transient levels. This allows the integration of electronic unitsof the CMS detectors in a safe way, such that will define future EMC standards
required by HEP community.
Due to the complexity of the CMS experiment, it has not been feasible to analyzeeach subsystem in detail; nevertheless general rules emerging from particularsubsystems, as the Hadron calorimeter (HCAL) and the Tracker, can be easily
extended to the rest of the sub-detectors. The EMC study has been focused on the
power supply distribution system, as CMS can be considered an isolated systemwith a sole galvanic connection done through the power supply system. For that
purpose, a simplified scheme of the power supply system has been used in thisresearch work to address EMC studies. Basically each sub-detector can be definedby these three constitutive elements:• Front end electronic (FEE)• Power supply (PS)• Cables (C)The FEE considers not only the amplification and the analogue to digital conversion
stages but also includes the particle detector device itself. This complete equipment
is the most sensitive part and has to be characterized by the susceptibility to
conductive noise and transients. Power supplies are characterized by the EM
emission, in particular, switching power supplies which generate high frequency
noise that can be coupled to the sensitive electronics via radiation or conduction.
Appropriate filtering and shielding may reduce the noise emission to levels that canbe tolerated by the FEE. Cables in general are the conductive link to couple noise
among sub-systems. The state of the art of technology nowadays allows reducing
the cable network because almost all high-speed digital communication between
the FEE and the acquisition system is performed via optical links. This fact highly
improves the immunity level of FEE against interferences, making easier thecomplete integration of such a large detector. Nevertheless, the study of thecommon mode and differential mode noise effects in long power cables constitutes
an important issue. These effects are crucial for the noise transmission from thepower supplies to the FEE and have to be taken into account in the selection of thecable and the design of the cable layout in the detector.

6Introduction

In the doctoral work planning, the thesis development has been divided in threemain parts, which correspond roughly to each of the 3 years of the Ph.D. During thefirst year, the study has been focused on the power supply system for CMS
sub-detectors, in particular HCAL and Tracker, and on the electronic layout of thesesub-systems. Using the Hadron calorimeter sub-system as experimental set-up,
tests of conducted noise generated by DC-DC converters have been performed.
Based on the obtained results, EMC filters have been evaluated to achieve a low
noise distribution system.
In the case of the Tracker sub-detector, it is planned to supply low voltage between
the control room and the detector (100-120 m.) using special multi-conductor cables
with different sections having different properties. A characterization of thesecables has been conducted based on cable models and measurements to define thecharacteristic parameters. The target is to define appropriated models to allow theanalysis of the stability of the power distribution system, the response to load
current perturbations and propagation of high frequency noise between the power
supply and the sensitive front-end electronics.
During the second year, studies have been focused around the development and theimplementation of the EMC tests, which will form the base line of the EMC
proposal. Tests were designed based on recommendations given by EMC engineersof the European Space Agency (ESA). A set of measurements was conducted during
the end of this year on CMS HCAL front-end electronics prototypes at FERMILAB,
USA.
The front-end electronics of the CMS Very Forward Hadron Calorimeter (HF)
sub-detector presents a special connection between the detector and the electronics.
This connection has been used to study the noise susceptibility to common modenoise signals and external electromagnetic fields of the electronic system at thedesign stage.
A 400Hz AC system is proposed to distribute power to all CMS sub-detectors. It
distributes high voltage to the periphery of the detector using a three-phase system.
On the detector, step-down converters transform the AC high voltage toappropriated DC low voltages. The implications of this kind of distribution in the
CMS detector have been also examined.
Finally during the last year, the results from the EMC tests have been analyzed. The
tests had been carried out in December of the previous year at FERMILAB. The
susceptibility of the FEE to EMI and perturbations has been defined. Also during
this period the effects of high frequency noise propagation through the power
cables and their implications in the final specifications of CM and DM emissions of
the power supply have been analyzed. As a final step, a set of additional EMC tests
on the power supply have been carried out with the aim of completing thecharacterization of the power supplies and output filters.

7

Introduction

The thesis report is organized as follows• CMS detector description (Chapter 1)• Grounding issues (Chapter 2)• Conducted emissions and filtering of switching power supply (Chapter 3)• Noise propagation effects on cables (Chapter 4)• Immunity of the FEE at design stage (Chapter 5).
• Susceptibility test of the FEE (Chapter 6)• Transients on the power supply distribution system (Chapter 7)

8Chapter 1
CMS detector

The Compact Muon Solenoid (CMS) is one of the four high-energy physics
experiments under construction at CERN for the Large Hadron Collider (LHC)
accelerator. In general, high-energy physics (HEP) detectors identify particle
interactions and measure their energy using sensitive detection devices such as
silicon and pixel detectors, optical devices, wire chambers, etc. The CMS andATLAS calorimeters at LHC constitute the biggest detectors ever designed for HEP
experiments. A short description of the main part of CMS detector and sub-systems
is presented as well as some details about the new circular accelerator the LHC.
Special attention is paid to two sub-systems of the CMS detector, as the EMC plan
has been based on several preliminary studies of these subsystems. These systems
are the Hadron Calorimeter (HCAL) and the Silicon Tracker Detector.

1.1

The Large Hadron Collider (LHC)The Large Hadron Collider (LHC) [1] is under construction at CERN laboratory on
the French-Swiss border west of Geneva, at the foot of the Jura mountains, in frontof the Alps. Combined with the LHC are 4 experiments called CMS [2], ATLAS [3],LHCb [4], and Alice [5].
The LHC consist on a 27 km circular accelerator placed 100 meters under the
ground, containing two beam pipes, capable of accelerating counter rotating
bunches of protons with beam energies up to 7 Tev. The bunches are focused tocollide within the experiments with centre of mass energies up to 14 Tev.
High-energy LHC beams need high magnetic bending fields. To bend 7 TeV protons
around the ring, the LHC must be able to produce magnetic fields of 8.36 Tesla bysuperconducting magnets.
LHC superconducting magnets will be operated at 1.9 K and will be capable tocarry currents of 15,000 amps and withstand forces, which build up to hundreds oftons per meter in the coils as the field rises. The LHC will be made of 5,000 LHC
superconducting magnets, which need a complex cryogenics system. In all, LHC
9

1. CMS detector

cryogenics will need 40,000 leak-tight pipe junctions, 12 million liters of liquid
nitrogen will be vaporized during the initial cool down of 31,000 tons of material
and the total inventory of liquid helium will be 700,000 liters. Figure 1.1 shows thelayout of the LHC.

Figure 1.1: LHC layout.

1.2 The Compact Muon Solenoid (CMS)
experiment
The goal of the CMS detector is to identify particles and measure their position
momenta or energy. An illustration of the CMS detector showing the various
sub-detectors and the super-conducting solenoid magnet is depicted in figure 1.2,
while figure 1.3 shows the CMS detector cavern. The detector is organized in
sub-systems as follows, from the interaction point to the periphery:10

1.2. The Compact Muon Solenoid (CMS) experiment

1. Tracker - (Silicon & Pixel detector)
The tracker is designed to measure precisely the momentum of the charged
particles. It is configured by a highly segmented and layered charged particle
detector involving millions of channels. Silicon pixel and strip sensors are used. The
presence or not of a charged particle is recorded in each channel allowing the tracks
to be reconstructed. The radius of curvature of the tracks in the magnetic fieldreveals the momentum of the particle.
2. Calorimeters
Electrons, protons and hadrons will be stopped by the calorimeters allowing themeasurement of their energy. The first calorimeter layer is designed to measure theenergies of electrons and photons with a high precision. Since these particles
interact electromagnetically, it is called the electromagnetic calorimeter (ECAL).
Particles, which interact via the strong interaction (hadrons), deposit most of theirenergy in the next layer, the hadronic calorimeter (HCAL). Neutrinos escape direct
detection but their presence can be inferred as an apparent energy imbalance after
the collision.
a. Electromagnetic Calorimeter (ECAL).
The electromagnetic calorimeter measures the energy of photons and electrons.
Almost no photon nor electron originating from proton-proton interaction point
pass beyond the ECAL.
b. Hadron Calorimeter (HCAL).
The hadronic calorimeter measures the energy of particles such as protons, pions,
neutrons and kaons. Almost no hadrons generated from proton-proton interactions
pass beyond the HCAL.
3. Superconducting Solenoid.
A long superconducting solenoid of 13m length and inner diameter 6m has beenchosen to create a uniform magnetic field of 4 Teslas in the direction of the beamaxis.
4. Muon detector.
The only particles generated from the proton-proton interactions to reach the muon
detector are muons and neutrinos. The muons are detected directly but theneutrinos escape undetected. The presence of a neutrino is inferred by missing
transverse energy.

111. CMS detector

Figure 1.2: Complete layout of the CMS detector.

121.3. Tracker Sub-system

Counting room
Optical channel
CMS detector

Cable channels
Balconies
Figure 1.3: View of CMS detector cavern.

1.3 Tracker Sub-system
The tracker detector [6] is of the utmost importance for an experiment designed toaddress the full range of physics. The CMS tracking system is designed toreconstruct tracks of high-p muons, isolated electrons and hadrons with a highmomentum resolution and a high efficiency. It is also designed to allow theidentification of tracks coming from detached vertices.
The tracker system is embedded in a 4 Tesla magnetic field, which forces low-p
particles into small radius helical tracks, resulting in reduced occupancy at larger
radius. The presence or not of charged particles is recorded in each channel and thetracks reconstructed. The curvature radius of the tracks in the magnetic field reveals
the momentum of the particle.
The layout of the tracker (Silicon & Pixel detector) [7] is shown in figure 1.4. This
volume operates at a temperature around -20 degrees.

131. CMS detector

Figure 1.4: View of the Silicon and Pixel detector.

1.3.1 Pixel detector
At the smallest section of beam line (around 10 cm radius), the interaction region of
the detector is surrounded by three layers of silicon pixel detectors. Two end-cap
disks cover the distance from 6 to 15 cm of radius. A mechanical scheme is depicted
in figure 1.5.

Figure 1.5: Pixel detector and pixel sensor.
The pixel layers are composed by modular detector units. Each module consists of athin, segmented sensor plate with highly integrated readout chips connected to14

1.3. Tracker Sub-systemthem using the bump bonding technique. The cell size in the pixel detector is
150x150 µ m and the total number of the chips of the pixel detector is around6
16000,which process the data from 45 ×10 pixels. Around 224 PS supplies thepower to the FEE of pixel detector. These voltages are 2.5 V and 1.8 V and the
maximum current per line will be less than 8 A. Also the pixel detector needsaround 112 power supply units to supply the HV (+600V). These units deliver only
a few mA.

1.3.2 Silicon detector
It is placed around the pixel detector. The cylindrical layers contain modules basedon rectangular detectors and the end-cap disk carries up to four rings ofwedged-shaped detector modules. Figure 1.6 depicts the geometry of the silicondetector.

Figure 1.6: Geometry of the Silicon detector. It shows a quarter of the detector.
The detector module is the basic functional component of the silicon tracking
system. Each module consists of three main elements:• Single side or double side silicon sensors.
• Mechanical support (Carbon fiber frame).
• Readout hybrids Front end electronics.
A general layout of a module is shown in figure 1.7.

15

1. CMS detector

Figure 1.7: Silicon detector module layout.
These modules are grouped in leaders and petals, as it is shown in figure 1.8 andfigure 1.9. Finally these leaders are mounted on two-structure wheel as it is shownin figure 1.4.
FEE
Sensors

Figure 1.8: Leader structure.

Figure 1.9: Petal structure.

16

1.3. Tracker Sub-system

1.3.3 Tracker Readout Electronics
A general block diagram of the front end electronic for the tracker [6] detector isshown in figure 1.10. The system processes analogue signals from 10 million
channels of micro-strip tracker with minimal power and material. Electronic noise
should be sufficiently low to ensure high efficiency and bunch crossing
identification. Performance is ultimately limited by the speed and magnitude of
detector signals and the radiation environment. It has four different parts; thedetector or sensor, the APVs [8] and the control unit located in the central part of thecalorimeter, and the front end controller and front end driver located in the
counting room.

DCU

APV
256:1

Opticaltransmitter

APV
MUX

PLL

Detector
CLK

T1
I2C

CCU
Tx/Rx

analogue
Front End Module opticallink

Control
moduledigitaloptical
link

ADC

FPGA
FPGA

TTCrx

RAM

Tx/Rx

TTCrx

Front End DriverµP

Front End Controller

Figure 1.10: General block diagram of the readout system of the silicon detector.
1.3.3.1 APV Front End circuit.
Each micro-strip is read out by a charge sensitive amplifier whose output voltage issampled at the LHC 40 MHz rate. Samples are stored in an analogue pipe line for a
few ms and following a trigger, are processed by an analogue circuit using aweighted sum algorithm to measure signal amplitude and associate the bunch
crossing with the hit. Pulse height data is multiplexed from a couple of front end
chips, sending differential signals over a short twisted pair cable to a laser, where171. CMS detector

electrical signals are converted to infrared and transmitted over 100 meter fibre
optic cable to the counting room adjacent to the cavern.
1.3.3.2 Optical Link
The optical link [9] employs edge-emitting semiconductor laser transmitters
operating at the telecommunications wavelength of 1310nm. Those lasers areassembled with single mode fibre pigtails in a low mass package developed incollaboration with industry. All link components have been shown to be extremely
radiation hard. Minimum current is required before laser actions starts. Abovethreshold the light output power is extremely linear with the drive current.
1.3.3.3 Front End Driver
Pulse height data are received by a photodiode - amplifier and the front end driverwhich digitizes and processes the signals, including reordering and pedestal
subtraction, and stores results in a local memory for the higher level dataacquisition.
1.3.3.4 Control System
The front-end controller supervises and controls the front-end electronics and also it
is the interface with the CMS Timing Trigger and command system. Digital opticallinks, using the same analogue link components, transmit triggers, clocks andcontrol data. Internally digital transitions are recovered by photodiode-amplifiersand distributed electrically by a communication and control unit (CCU) to detector
modules. Clocks are recovered by phase locked loop (PLL) chips on each module
for high reliability and minimum phase jitter. CCU modules can be configured asrings to match the tracker topology and reduce cost
1.3.3.5 Low and High voltage PS distribution system.
Tracker sub-detector [10][11] has proposed several topologies to power-up thefront-end electronics. In such proposals, the primary 400Hz AC distribution system
is rectified and later, DC-DC converters transform with high efficiency the 48 V into
the LV and HV required by the front-end (FE) electronics, which are locallyregulated using sense wires. The final location of these converters is not yet defined,
one proposal planned to locate the converters between 20-40 mts. from thefront-end electronics. The other plan is to place both the rectifiers and the DC-DC
converters in the counting room (100 or 120 meters far from the detector) due to theintense magnetic field that exists inside the detector.

18

1.3. Tracker Sub-system

The total number of tracker detector modules is 13884 and the total number of
APV's is around 80000, (i.e 100 million of read-out channels.). Between 12 and 6
modules are grouped together to create a PS group, each of them is supplied by a PS
unit. Tracker needs around 250 kW in order to work. About 2000 power supply
units power the silicon tracker detector. Each of them supplies three different
voltages (Table 1.1) to a specific group of modules.
Voltage (V)Max. current
(A)

LV -1

+ 2.5 V10

LV -2

+ 1.25

8

HV

+ 600

0.02

Table 1.1 Tracker power supply values per powerchannel.
Two different long and complex cables are used to supply the power from the
DC-DC converters to the tracker FEE. These cables are split in three or twosegments. The length as well as the type of cable is different for each segment.
Basically three types of cable can be found in tracker sub-detector.
-

Low impedance cable (LI)-

Multiservice cable (MSC) - Copper [12]-Multiservice cable (MSC) - Aluminum

These complex cables have HV, LV, signal and control wires located inside analuminum and copper braid shield. Figure 1.11 shows the schematics of the LIC and
MSC copper cable.

ab

Figure 1.11: Tracker power cables a) MSC cable and b) LIC cable.

191. CMS detector

1.4 Electromagnetic Calorimeter (ECAL)The CMS electromagnetic calorimeter [13] is composed by scintillating crystals andoffers excellent performance for energy resolution since almost all of energy ofelectrons and photons is deposited within the crystal volume. CMS has chosen leadtungstate crystals, which have high density, a small Moliere radius and a shortradiation length allowing a very compact calorimeter system.
The electromagnetic calorimeter system is placed around tracker detector in the 4
Tesla magnetic field. Electrons and protons will be stopped by the calorimeters
allowing their energy to be measured. There are two different parts on thissub-detector:• Electromagnetic calorimeter.
• Pre-shower detector.
The layout and some details of the ECAL system are shown in figure 1.12.

Modules
Crystals

Pre-shower

Figure 1.12: Interior view of the Electromagnetic calorimeter and Pre-shower detector.
The CMS electromagnetic calorimeter consists of over 76000 lead-tungsten crystals
equipped with avalanche photodiode (APD) or photodiodes and associatedelectronics operating in a challenging environment, with a magnetic field of 4 Tesla,high radiation dose and difficult access for maintenance.
The detector module is the basic functional submits of the ECAL system. Each
module consists of three elements.
• Crystal• Basket or mechanical support

20

1.4. Electromagnetic Calorimeter (ECAL)• Readout electronics (FEE)
A general layout of a module is shown in figure 1.13:

Figure 1.13: General layout of a ECAL module.

1.4.1 ECAL Read out electronics
A general block diagram of the front end electronic for the ECAL detector [14][15] isin figure 1.14. The first element in the front-end chain is the lead- tungsten crystal,
which converts energy into light. The light is converted into a photocurrent by the
APD. The relative low electric current delivered by the photodetector device isamplified, shaped and converted to a voltage waveform. This signal is thenacquired and digitized. The resulting data is transmitted off the detector via opticalfibre to the read out located in the counting room.

Figure 1.14: Scheme of the ECAL readout system.
The upper level read-out has four main functions:21

1. CMS detector• Formation of trigger tower energy sums
• Pipelining (Storing the data until arrive of level -1 trigger decision)• Transmission of the data of the triggered event to the data acquisition system• Providing interface functions for the 'on detector electronics'.
ECAL needs around 450 kW in order to work. About 750 power supply units [16]bias the ECAL front-end electronics. Each of them supplies 4.5 V with a maximum
current of 100 A. This voltage is used to supply the power to a low voltage regulator
board where LV regulators transform the input voltage into the final one required
for the electronics, which is 2.5 V.

1.4.2 Pre-shower detector
CMS utilizes a pre-shower detector [13] in the endcap region. This region is shownin figures 1.12 and 1.15. Its main function is to provide a separation between two
different particle decays ( γ – π 0 ). The CMS pre-shower consists of two leadradiators having about 2 and 1 radiation length thick respectively, each one
followed by a layer of silicon micro-strip detectors. Both layers of detectors havetheir strips orthogonal to each other. The first layer has vertical strips and measures
the horizontal position of shower particles, whilst the second layer has horizontal
strips to measure the vertical position of particles. The energy deposited in thesilicon detectors is proportional to the energy deposited in the lead, so it can beused to apply a correction.
The pre-shower has a cylindrical shape about 18cm long (along the beam axis), theinner radius is approximately 45cm whilst the outer radius is about 120cm (h =1.65).
The total area to be covered by silicon detectors is about 16.4 square meters(Around half of the one used by the Tracker in CMS).

Figure 1.15: Location of the Pre-Shower in the end cap.

22

1.4. Electromagnetic Calorimeter (ECAL)

The silicon detectors used in the pre-shower do not have to present ultra-finegranularity like in the Tracker detector. A strip pitch around 2mm is sufficient forthis application. To reduce the number of channels to a minimum, it has beenchosen the largest detectors that are easily integrated.
The motherboard is the basic functional submit of the Pre-shower detector. Each
module consists of three elements.
• Single side or double side detectors• Mechanical support• Readout hybrids (Front end electronics)
This module is shown in figure 1.16.

Figure 1.16: Pre-shower module layout.
1.4.2.1 Pre-shower read-out electronics
The basic requirements for the pre-shower electronics are: 1) the front-end musthave a large dynamic range (due to the large fluctuation in particle energies to bemeasured), 2) low noise, and 3) a "pipeline memory" to store signals until a trigger
signal arrives. The CMS ECAL and Tracker detectors require similar electronics, sosome common parts have been adapted to the pre-shower needs. However, someparts for the front end electronics of the Pre-shower are such as the 'PACE' circuit[17] (Pre-shower Analog CMS Electronics). The PACE has 32 channels (one for eachof the micro-strips on a single silicon detector) with 160 memory cells for eachchannel. The analog signals from the PACE are fed into an analog-to-digitalconverter (ADC) with a large dynamic range. A suitable ADC is the CRIAD, also23

1. CMS detector

designed at CERN. The output digital signals are sent via optical cables to VME
crates, located in the counting room. The VME electronics performs
zero-suppression and some signal processing before feeding the data to the CMS
data acquisition system (DAQ). The motherboard (MB) is located into the detector,
while the DDU is located into the counting room.
Pre-shower needs around 21 kW in order to work. About 48 power supply units
power the pre-shower detector. Each of them supplies the 4.5 V voltage required at
the input of the FEE, where LV regulators transform the 4.5 V in the final low
voltage required by the analog and digital part of the FEE.

1.5 MUON chambers sub-system
Muons are expected to provide clean signatures for a wide range of physics
processes. The task of the muon system [18] is to identify muons and provide, in
association with the tracker, a precise measurement of their momentum. In
addition, the system provides fast information for triggering purpose. The muon
detectors, placed behind the calorimeters and the coil, consist of four muon stations
interleaved with the iron return yoke plates. They are arranged in concentric
cylinders around the beam line in the barrel region, and in disks perpendicular tothe beam line in the end caps. They are shown in light blue in figure 1.17.

Figure 1.17: Layout of barrel muon disk (Blue colour).
CMS will use three types of gaseous particle detectors for muon identification: 1)
Drift tubes (DT) in the central barrel region, 2) Cathode Strip Chambers (CSC) in theend cap region, and 3) Resistive Parallel Plate Chambers (RPC) in both the barrel

24

1.5. MUON chambers sub-systemand end caps. The DT and CSC detectors are used to obtain a precise measurementof the position and thus the momentum of the muons, whereas the RPC chambers
are dedicated to provide fast information for the Level-1 trigger.
1. Drift tubes.
Drift tubes are used in the barrel where the magnetic field is guided and almost
fully trapped by the iron plates of the magnet yoke. Each tube contains a wire withlarge pitch (4 cm), and the tubes are arranged in layers. Only signals from the wires
are recorded - resulting in a moderate number of electronics channels needed toread out the detectors. When an ionizing particle passes through the tube, it
liberates electrons, which move along the field lines to the wire, which is at positive
potential. The coordinate on the plane perpendicular to the wire is obtained withhigh precision from the time taken by the ionization electrons to migrate to the wire.
This time (measured with a precision of 1ns), multiplied by the electron drift
velocity in the gas, translates to the distance from the wire.
A DT layer is put together by gluing to an aluminum plate a set of parallel
aluminum beams. The wires are stretched and held by the appropriate end plugs,
and the layer is closed by another aluminum plate. Groups of four layers are grownin this way on a precision table. Copper strips are previously glued to thealuminum plates in front of the wire to shape better the electrostatic field. The
largest DT chamber to be used in CMS will be 4 m x 2.5 m.
2. Cathode Strip Chambers.
CSCs are used in the end cap regions where the magnetic field is very intensive andvery inhomogeneous. CSCs are multi-wire proportional chambers in which one
cathode plane is segmented into strips running across wires. As avalanche isdeveloped a wire induces a charge on several strips of the cathode plane. In a CSC
plane, two coordinates per plane are made available by the simultaneous andindependent detection of the signal induced by the same track on both the wires
and the strips. The wires give the radial co-ordinate whereas the strip measures theangle. In addition to providing precise space and time information, the closely
spaced wires make the CSC a fast detector suitable for triggering. The CSC modules
containing six layers provide both robust pattern recognition for rejection of
non-muon backgrounds and also efficient matching of external muon tracks tointernal track segments.
Six layers CSC are built assembling together 7 Honeycomb panels. Three of them
support two wire planes each, one on each face of the plate, wire at the same time.
The other four plates have the etched strips. The two inner plates have strips on
both faces, whereas the two outer plates (closing the chamber) have strips only on
one face.
3. Resistive Parallel Plate Chambers.

25

1. CMS detector

RPCs are fast, gaseous detectors whose information is the base of the triggering
process. RPCs combine good spatial resolution with time resolution of 1 ns,
comparable to that of scintillators. The RPC is a parallel plate counter with twoelectrodes. It is made of plastic material with very high resistivity. This allows theconstruction and operation of very large and thin detectors that can operate at highrate with high gas gain. Resistive Parallel plate Chambers are fast gaseous detectors
whose information is the base of the triggering process. The total number ofchamber of the Muon system is around 1800.

1.5.1 Muon Sub-system read-out electronics
1. FEE - Cathode Strip Chambers.
Cathode and anode front-end boards (FEB) are mounted directly on the CSC
modules. The data from the FEB are sent by cable to the readout motherboard, also
mounted on the chamber. The motherboard serves as link between FEBs and therest of the detector. It sends both the readout and trigger data to the central DAQ
system and level-1 trigger system located into the counting room. It also receives
the trigger, timing and control (TTC) signals from the counting room and distributes
them to the FEBs. This connection is implemented using an optical link. A generallayout of the readout system of CSC is shown in figure 1.18.

Figure 1.18: Readout structure of CSC.
2. FEE - Drift tubes.
The FEE is basically composed of both the read-out boards (ROB) and the read-out
servers (ROS). All these units are housed in crates and are placed in racks on the26

1.5. MUON chambers sub-systembalconies (periphery of the detector). The FEE, placed inside the gas volume, will
provide discriminated differential signals that are transmitted to the read-out
boards (ROB) using a 25-m twisted-pair cable.
3. FEE - Resistive Parallel Plate Chambers.
A general scheme of the front-end control system and the data transmissioninterface to the optical link is shown in figure 1.19. The data from the front-end
board (FEB) is sent, using flat cables, to a link board where it is processed andtransmitted to the trigger processors. The FEBs are located close to the detector andare connected via an optical link, with both the RCP trigger crates and the network.
It also supplies the LHC clock and broadcast test pulses to the FEB.

Figure 1.19: Layout of the front-end system and transmission interference.
Muon chambers need to operate about 250 kW and it will be provided by 3000
power supply units with different voltages and currents. A short summary ofvoltages and currents per sub-system are shown in table 1.2.
Voltage

Max Current

HV - RPC

10 kV

0.2 mA

HV - CSC

5 kV

0.2 mA

HV - DT4 kV

0.1 mA

LV-1

2.5 V

10 A

LV-2

3.5 V

42 A

LV-3

5.0 V

38 A

LV-4

6.5 V

56 A

LV-5

7 V

13.5 A

Table 1.2 Power requirements of Muon chambers.

27

1. CMS detector

1.6 Super-conducting Magnet Sub-system
The CMS magnet sub-system [19] consists on a super-conducting coil, the magnet
yoke (barrel and end cap), a vacuum tank and ancillaries such as cryogenics, power
supplies and process controls. The main parameters of this sub-system are: amagnetic field of 4 Tesla, a yoke diameter of 14m across flats, an axial yoke length
including end caps of 21.6m and a total weight of about 11000 tons. It will be thelargest super-conducting magnet system in the world. The energy stored into it, if
liberated, will be large enough to melt 18 tons of gold. A general view of the magnet
is depicted in figure 1.20a.

ab

Figure 1.20: 4 Tesla CMS Magnet system.
The magnetic flux generated by the super-conducting coil is returned via a 1.5m
thick saturated iron yoke (figure 1.20b). The yoke is divided into the barrel and theend cap yoke. The barrel yoke is a 12-sided cylindrical structure, divided into fiverings. It is 13.2 m long, giving a total iron mass for the barrel of about 7000 tons. The
central barrel ring, centered on the interaction point, supports the super-conducting
coil. Each barrel ring is made up of three iron layers. Connecting brackets join
together the steel plates forming the three layers and provide the required structural
rigidity. The central barrel ring is the only stationary part around the interaction
point and it is used to support the vacuum tank and the super-conducting coil. The
other four-barrel rings and the end-cap disks slide on common floor rails, runningin the beam direction, to allow insertion and maintenance of the muon stations.

28

1.6. Super-conducting Magnet Sub-system

1.6.1 Super-conducting coil
The super-conducting coil [20] consists of the coil itself and the ancillarysub-systems required for its operation and protection. The structure of thesuper-conducting coil is self-supporting, whereby magnetic forces are resisted
where they are produced. The reinforced conductor of the four-layer CMS coil cansustain by itself all the induced magnetic forces.
Bellow some details of the magnet are shown.

Magnetic induction at interation point

4 TeslaMagnetomotive force

42.20 MaT

Conductor peak magnetic field

4.6 T

Winding overall current density

A
12.68 ----------2mm

Stored energy

2.69 GJ

Magnetic radial pressure

6.4 Mpa

Axial compressive force at midplane

148 MN

Operating current

19.5 kA

Inductance

14.15 H

Total number of turns

2168

Turns per layer

542

Dump resistor

0.05 Ω

Dump voltage

1000 V

Dump time constant

283 s

Table 1.3 Magnet characteristics.

1.6.2 Power supply system of Super-conducting solenoid
A 2-quadrant thyristor power converter deliver a coil current of 20 kA at a
maximum ramping voltage of 16 V. The current ramping time necessary to power
up the magnet will be some hours.

29

1. CMS detector

The magnet can be discharged by different modes that vary from hours in normal
operation up to around ten minutes during emergencies. In case of a main power
failure, electrical power could be generated for helium recovery by using some ofthe stored magnetic energy.

1.7 HCAL sub-system
1.7.1 System generalities
The Hadronic Calorimeter (HCAL) [21] plays an essential role in the identificationand measurement of quarks, gluons, and neutrinos by measuring the energy anddirection of jets and of missing transverse energy flow in events. Missing energy
forms a crucial signature of new particles, like the super-symmetric partners ofquarks and gluons. For good missing energy resolution, a good calorimetry is
required. The HCAL will also aid in the identification of electrons, photons andmuons in conjunction with the tracker, electromagnetic calorimeter, and muon
systems.

HCAL Forward

HCAL Barrel

HCAL Endcap
Figure 1.21: One-quarter view of the CMS detector showing the location of HCAL
sub-system (in Yellow).
The hadron barrel (HB) and hadron end cap (HE) calorimeters are sampling
calorimeters with 50-mm thick copper absorber plates interleaved with 4-mm thickscintillator sheets. Figure 1.21 depicts a view of the detector showing the location of
the HCAL parts. Copper has been selected as the absorber material because of its

30

1.7. HCAL sub-systemdensity. The HB is constructed of two half-barrels, each of 4.3 meter length. The HE
consists of two large structures, situated at each end of the barrel detector andwithin the region of high magnetic field. Because the HCAL barrel inside the coil is
not sufficiently thick to contain all the energy of high-energy showers, additional
scintillation layers (HOB) are placed just outside the magnet coil. The full depth ofthe combined HB and HOB detectors is approximately 11-absorption length.
There are two hadronic forward (HF) calorimeters, one located at each end of the
CMS detector, which complete the HCAL coverage. The HF detectors are situated ina harsh radiation field and cannot be constructed of conventional scintillator andwave-shifter materials. Instead, the HF is built of steel absorber plates; steel suffers
less activation under irradiation than copper. Hadronic showers are sampled atvarious depths by radiation-resistant quartz fibers, of selected lengths, which are
inserted into the absorber plates.

1.7.2 HCAL FEE
1.7.2.1 Introduction
The readout of the CMS hadron calorimeter is built through a chain of systemelements beginning with photo-detectors coupled to the light produced in thecalorimeter detection media and ending with memory storage of digitized results inan on-line processor farm. Electronics systems [22][23] are located in three different
areas on the CMS experiment complex:• In the control room at grade level 150 meters above the accelerator tunnels andcaverns.
• In the shielded underground service room about 100 meters inward from thebeam line (USC55).
• On or adjacent to the detector in the underground cavern centered on the beamline.
Signal processing functions required from the calorimeter readout electronics chain
during colliding beam operations can be summarized as follows:• Analogue signal conditioning of photo-detector responses• Digitalization of conditioned analog signals at the beam-crossing rate of 40MHz.
• Transmission of digitized values from the detector to the control room using 1.6
Gbit optical links.
• Linearization and conversion of the front-end results into deposited energy
values at 40MHz.
• Generation and transmission of filter-extracted first level trigger information at
40MHz.

31

1. CMS detector• Pipeline storage of linearized energy values during the first level trigger
decision interval at 40MHz.
• Buffering the linearized time samples at the average first-level trigger accept rateof 100 kHz• Generation of the second level trigger information at 100 kHz.
• Formatting, organising and transferring of trigger and linearized time sample
data to the event builder at 100 kHz.
All of the 40MHz signal processing operations at the very front end of the system
are synchronous with the accelerator operations and are phase locked to the beamcrossing. The higher levels of the readout system operate at an average "interesting
event" rate of 100 kHz and are de-coupled from the synchronous, pipelined front
ends by a set of de-randomising buffers. Other modes of operation of the readout
system, e.g. data acquisition for the light flasher calibration system ordetermination of pedestal values are far less demanding.
1.7.2.2 System configuration
Partitioning of the readout chain is subject to several important global and spaceconstraints, and is impacted strongly by considerations of maintainability. Access tothe underground cavern is limited, regions along the sidewalls can be accessed on atwice a month basis. For the inner part of the detector access could be possible on a
yearly basis with difficulty. The CMS first level trigger system can not be located on
the surface because of the time delays introduced; it is located as close to thedetector as possible in the adjacent underground service room. Sufficient radiation
shielding is planed so that this area can be accessed during accelerator operation.
The remainder of the data acquisition and trigger system is located on the surface
due to the limited size of the shielded underground service room.
The HCAL sub-detector has electronics systems located on:• The detector.
• The side walls of the collision cavern and the periphery of the detector.
• The underground service room or counting room.
• The surface.
Analogue signal conditioners and digitization electronics are located at thephoto-detectors attached to the calorimeter elements; trigger and data acquisition
electronics are located in the underground service room; and readout control,
formatting, and interfaces to higher levels of the system are located on the surface.
1.6Gbit/s optical links connect the front-end electronics on the detector to thetrigger and data acquisition systems in the counting room. Standard
communication links bring the digitized data to the surface at the reduced rate
corresponding to the first level trigger accepts. Low voltage power supplies are32

1.7. HCAL sub-system

located on the periphery of the detector and other utilities are placed along the
sidewalls of the cavern. Figure 1.22 shows a block diagram of the front-end
electronics and the data acquisition system. (Note: The optical link depicted in thisfigure is 800Mbits/s but recent tests have shown good transmission performance ofthe link operating at 1.6Gbit/s. This speed has picked for the final version and thenumber of TX circuits in the FE readout module will be reduced to two).

Figure 1.22: FEE and DAQ system layout.
1.7.2.3 Front-end electronics
The front-end electronics system comprises those components located on thedetector in close proximity to the calorimeter photo-detectors. These components
provide the functions of analog signal conditioning, digitization, synchronization/
control and data transmission. Performance and reliability risks associated withplacing the digitizer portion of the readout electronics directly on the calorimeter,where routine service is problematic, have been evaluated and incorporated into therequirements for reliability.
A functional block diagram of the system is shown in figure 1.23. Analogue signal
conditioning is done using a multi-range current splitter and gated integrator, theQIE (Q for charge, I for integrator, and E for range encoding) ASIC [24]. The lastversion designed for CMS HCAL incorporates an input difference amplifier and anon-linear 5 bit flash ADC. The outputs of this ASIC are 2 bits of range information
and a 5 bit signal corresponding to the FADC outputs. This result corresponds to a 7
bit pseudo floating-point format with 2 bits of range (or exponent) and 5 bits ofcharge (or mantissa). A separated digital ASIC, denominated CCA, is needed to33

1. CMS detector

control and synchronise the QIE channels with the accelerator radio frequency time
structure. Parallel digitized information is serialised using another especiallydesigned ASIC, denominated GOL [25].

Figure 1.23: Block diagram of the front-end card.
1.7.2.4 Analogue signal conditioning
In the central region (Barrel/ End-Caps), the calorimeter detection elements are
scintillators readout with wavelength shifting plastic fibres and the photo-detectors
are hybrid photodiodes (HPD). The shape of the light pulse produced is an initial
step followed by an exponential decay corresponding to the fluorescencecharacteristics of the combined scintillator-waveshifter system. This time constant,using a single exponential approximation, has been measured to be 11.3 ns. Thus
the expectation is that 89.1% of the light signal occurs in the 25 ns, 9.9% occurs in thenext 25ns interval, and 1.0% occurs in the third 25 ns interval on average. The signal
is stretched further, when the 5 to 10 ns impulse response of the photo-detector.
Differences in optical path lengths of the elements of a tower are taken into accountand convolved with the light signal; 68% is in the first interval, 29% in the second,and 3% in the third on average. Finally, statistical fluctuations on these averagevalues are significant, especially for the case of low levels where only tens ofphotoelectrons are involved.
Since the signals from at least two following beams crossing after the one of interest
must be added to obtain the true value, the current produced by the photo-detector
must be integrated and digitized over each 25ns interval separately. This is the casewhen there is pile up of signals from adjacent crossing as estimating the baseline
shift requires charge integral results for several early crossing. Therefore, the central
systems require a gated integrator that is reset every crossing, has a precise aperture34

1.7. HCAL sub-system

as close to 25 ns as practical, and suffers negligible charge loss at each 25 ns
boundary.
In the forward region, the calorimeter detection elements are quartz fibres directly
viewed, and the photo-detectors are photo-multipliers tubes (PMT). The signal
produced is due to Cerenkov light from relativistic shower particles, so that it isvery fast. Test beam results with small, fast phototubes indicate that the entire
signal can easily be made to occur in less than 25 ns. Pile up of signal from adjacentcrossing thus does not occur, and the pedestal (or the baseline) depends only on theamount of out-of-time background.
Given that the signal duration is smaller than the bunch-crossing interval, a gatedintegrator which is reset each crossing is also appropriated for the forward
calorimeter. The requirement on aperture precision is modest, 10% is adequate, andthe issue of charge loss at the 25ns boundaries is not relevant to the principal
measurement.
1.7.2.5 Dynamic range
The main compartments of the barrel and end cap calorimeters, HB and HE, arerequired to respond reliably to minimum ionising particles in order to contribute to
the muon trigger and for off-line identification of muons. This situation determines
the details of the low end of the dynamic range. At the high end, studies haveshown that the largest energy deposition expected in a central or endcap
calorimeter main compartment over 10 years of operation at full luminosity is of theorder of half the bean energy, or 3.5TeV. Actual jet energies could be higher, but theenergy in a jet is spread over several towers spatially and is shared between theelectromagnetic and hadronic depth segmentations. In terms of photoelectrons,
these requirements correspond to a dynamic range of 1 to 35000 or 15 bits.
The outer compartments of the central and end cap calorimeters, HOB and HOE,
are required to be sensitive to minimum ionizing particles. However, the highest
energies seen are significantly lower than those in the main compartments due tothe depth at which these layers are located. The physics dynamic range necessary isfrom minimum ionizing to a rate 1TeV occurrence. When the photoelectron yieldand the low-end granularity considerations are taken into account, the requirement
is for a 16000 to 1 dynamic range or 14 bits.
For the forward calorimeter, the single photoelectron signal is extremely importantas it corresponds to more than a GeV of energy. Cerenkov light, even when directly
viewed, is quit weak in intensity. Test beam studies show than the calibration for themain compartment is approximately 0.4 photoelectrons per GeV. Allowing theupper end to extend to the full beam energy gives a dynamic range of 1 to 3000
photoelectrons, or slightly less than 12 bits. Providing an extra factor of 435

1. CMS detector

granularity for single photoelectron performance then gives an overall dynamic
range requirement between 13 and 14 bits.

1.7.3 FEE Components
1.7.3.1 Overview
The requirements for the front-end electronics detailed above call for a low noise,high frequency, wide dynamic range digitiser system. In particular the combinationof 4500e- rms noise, 1 fC least count (1fC/LSB1), 40 MHz clock frequency, and 16
bits dynamic range is unique to the LHC conditions. The LHC case, impose
additional special requirements on reliability tolerance, magnetic field immunity,
cross-talk and capabilities for DC current measurements.
The front-end electronics design is based on an improved version of the analogue
processing QIE ASIC development. As shown in figures 1.22 and 1.23, the building
blocks of each FEE module are the analog + digitiser ASIC chip, a control ASIC anda serializer /diode driver. Due to the extreme space constraints present in the barrel
and end cap regions, a modular 6 channels format has been selected as indicated infigure 1.23. Each component is described in the following subsection, putting moreemphasis on the analog section of the QIE. 2 channels detail of the FEE board is
depicted in figure 1.24.

Figure 1.24: FEE channel block diagram.
1 LSB=Less significant bit36

1.7. HCAL sub-system

1.7.3.2 Analogue conditioning and digitising ASIC-QIE
The QIE is a multi-range device, the input current is simultaneously integrated on
all the ranges, and comparators are used to select the lowest range that is not fullscale. The outputs are voltages representing the integrated charge plus a two-bit
gray code indicating the range. Operations are time multiplexed and pipelined toallow signals to settle and to make the reset interval the same as the integrationinterval. Latency is 100ns as the pipeline is four clock cycles deep.
The QIE contains multiple sets of eight capacitors of a uniform value. Attached tothis structure is a current splitter. Matched transistors in common baseconfiguration and connected in parallel will share the current driven through themequally. This property is exploited in the QIE design to apportion a fixed fraction of
the input current to a given capacitor in the array. Each capacitor receives a fractionof the current of its lower range neighbor. For example, if the splitting were simply
by powers of two, the capacitor of range n would receive 1/2n of the input current.
The basic circuitry inside the QIE is repeated by four to allow the pipeline
operations of a) integration, b) settling and holding, c) Analog to Digital conversion
and d) resetting. This means when one set of capacitors integrates the input current
for one beam crossing interval; others are holding a voltage proportional to theintegrated charge in the previous clock cycle, other set is being read-out by the
FADC while the fourth group is being reset. Due to chip has 4 sets of internal
capacitors, they have to be calibrated to improve the exactitude of the charge
measurement. These capacitors are calibrated before operation and their values are
stored in a look-up table resident in the counting room. The QIE identifies the set of
capacitors using during the integration by two bits called 'Cap ID' (Figure 1.24).
Summary of QIE requirements• 40MHz operation.
• 4 stage pipelined device (25nsec per stage)-

Charge collection-

Settling-

Readout (A to D conversion)

-

Reset

• Inverting and Non-inverting inputs. The inverting configuration is used at
Barrel and End-caps FEE, where the photo-detectors are hybrid photo detectors(HPDs). The non-inverting configuration is used at very forward FEE, where thephoto-detectors are photo-multipliers.
• The front-end amplifier has two inputs. Both currents are integrated and theresult is subtracted before the digital conversion. Equivalent to a ’Differential
amplifier'37

1. CMS detector• Impedance of both input terminals 50 ohms or 92 ohms.
• Internal non-linear Flash ADC• FADC Differential Non-Linearity < .05 LSBs
• Outputs: 5 bits mantissa, 2 bits range exponent and 2 bits Cap ID• 16 bits dynamic range• Charge sensitivity: Barrel / End-cap version (HPD photo-detector) = 1 fC/LSB• Calibration mode = 2000 electrons or 0.32 fC/LSB• Very Forward Cal. version (Photo-multiplier) = 2.7 fC/LSB• base thermal noise: 4500 electrons rms (Barrel / End-cap version).
• Systematic errors small enough to measure a few least count source current• Charge loss at sample boundaries less than 2%• Cross-talk less than 2%
1.7.3.3 Channel control ASCI (CCA)The CCA provides the following functions:• The processing and synchronization of data from two QIEs (Figure 1.24).
• The provision of phase-adjusted QIE clocking signals to run the QIE charge
integrator and Flash ADC.
• Checking of the accuracy of the Capacitor IDs, the Cap Ids from different QIEs
should be in synchronism.
• The ability to force the QIE to use a given range.
• The ability to set Pedestal DAC values.
• The ability to issue a test pulse trigger.
• The provision of event synchronization checks. A crossing counter will be
implemented and checked for accuracy with every beam turn marker.
• The ability to send a know pattern to the serial optic link.
• The ability to 'reset' the QIE at a known and determined time.
• The ability to send and report on any detected errors at a known and determinedtime.
1.7.3.4 Data links
The CMS trigger system needs information about every crossing. The data will besent 100-150 meters from the detector to the trigger system using either a 1.6 Gbit/s
or a 800Kbit/s optical serial link. An ASIC serializer and laser driver, denominated
GOL, is used to achieve the parallel to series conversion and transmission. This chip38

1.7. HCAL sub-system

has been designed at CERN and will be used for perform this function by different
CMS sub-detectors and also other LHC experiments. The principal characteristics ofthis device are:• Synchronous operation (constant latency)• Transmission speed-

Fast: 1.6 Gbps, 32 bit data input @ 40MHz.

-

Slow: 0.8 Gbps, 16 bit data input @ 40 MHz.

• Two encoding schemes-

G- Link-Fibre channel (8B/10B)• Interfaces for control and status registers
-

I2C

-

JTAG

1.7.3.5 Linear Rad-Tolerant Voltage Regulators
Each 6 channel board requires 3 power supplies, +5V, +3.3V and +2.5V. The two first
voltages are regulated using specially designed Rad-tolerant low-drop voltage
regulators. This design achieves an output/input differential mode filtering of morethan 40dB at a frequency of 1MHz. The +2.5V bias is generated from the +3.3V biasline using discrete transistors and operational amplifiers.

1.7.4 Front-end Electronics Readout Boxes (RBX).
In the case of the barrel and end caps units, the front-end electronics is close to the
HPD photo-detector and both are resident into readout boxes (RBX). For the very
forward calorimeter, the electronics is resident into 3U crate separated about 4 mfrom the photo multiplier tubes.
1.7.4.1 Channel distribution per RBX• Barrel Hadron (HB):-

18 + 18 RBXs installed in the detector.

-

4 HPDs per box, 12 boards per box.

-

72 channels per box, 3 boards per HPD.

• End Cap Hadron(HE):39

1. CMS detector-

18 + 18 RBXs installed in the detector.

-

4 HPDs per box, 12 boards per box.

-

72 channels per box, 3 boards per HPD.

• Outer Barrel/End Cap (HO) :-Wheel 0:a. 12 RBXs installed in the detector.
b. 3 HPDs per box, 9 boards per box.
c. 54 channels per box, 3 boards per HPD.
-

Wheels +1/-1:

a. 6 + 6 RBXs installed in the detector.
b. 4 HPDs per box, 12 boards per box.
c. 72 channels per box, 3 boards per HPD.
-

Wheels +2/-2: 6 + 6 RBXs installed in the detector:

a. 4 HPDs per box, 12 boards per box.
b. 72 channels per box, 3 boards per HPD.
• Very Forward Calorimeter:-

1242 + 1242 channels.

-

Boxes hold up to 30 PMTs.

1.7.4.2 RBX layout
All the equipment installed into and around the detector must be thermally neutral.
It forces to design a water-cooled system around the RBX to absorb the heatgenerated by the FEE. A specially designed aluminum frame with water-coolingpipes is used to hold the FEE boards, photo detectors and the auxiliary opticalsystem.
RBXs hold 4 read-out units. Each unit contains 3 FE boards connected to themulti-anode HPD and an optical decoder (ODU) where fibres coming from thescintillators are coupled to the HPDs. This read-out unit is an 'optical processor' in
the sense that light coming from the scintillators is electronically processed,
digitised and optically transmitted to the counting room. Figure 1.25 shows an
overview of the connection of the HPD with both the FEE board and the ODU. Each
unit is powered though an internal back-plane that also distributes 40 MHz clock to40

1.7. HCAL sub-system

the units, provides a path for RBX bus (serial communication bus) and temperaturemonitors.

Figure 1.25: Overview of the read-out module.
For the very forward HCAL, the FE boards are housed into standard 3U crates andtwo coaxial cables per channel connect the QIE inputs to the photo-multipliers(PMTs). This distance is about 4 m. Power and clock is distributed to each FE board
using a back-plane.
1.7.4.3 Power requirements per RBX
Each 6 channel board requires +6.5V and +4.5V as primary voltages [26]. As it waspresented above, these voltages are adjusted at board level by linear regulators to+5V, +3.3V and +2.5V. The currents required by each board are 0.55A for the +6.5V
supply and 0.77A for +4.5V supply.
Including some safety margin, the total design currents per RBX are the following:+ 6.5 V+ 4.5 V

Barrel6.77 A

14.6 AEnd Caps

9.42 A

18.5 A

Outer Hadron

5.18 A

12.24 A

W+1

6.77 A

14.6 A

W+2

6.77 A

14.6 A

Table 1.4 Power consumption.

41

1. CMS detector

1.8 PS distribution system of CMS detector
1.8.1 Introduction
The DC low-voltage (LV) and the DC high voltage (HV) distribution system in CMS
are based on a primary 400Hz AC distribution system [27]. Figure 1.26 shows theschematic of the power supply distribution of CMS detector. The 400Hz AC system
is proposed as primary power source to distribute power to all CMS sub-detectors.
It is composed by independent motor-generator sets for each sub-detector thatconverts the 50Hz AC mains to a 400Hz three-phase 400V system. The AC powersystem will distribute high voltage between the surface level and the counting roomor the periphery of the detector.

Figure 1.26: PS distribution block diagram of the CMS system
AC/DC converters located at both the periphery of the detector and the counting
room rectify the three phase of the 400Hz system and generate a primary DC
voltage of 48V for some sub-detectors and around 8V-15V for other sub-detectors.
For some sub-systems using the extra DC low voltage, AC/DC converter are
located in the periphery of the detector and the low voltage is distributed to the FEE
units, where on board LV regulators set the final voltage required by the detector
electronics. Other sub-systems needs the DC voltage (48 V) from the rectifiers to
supply intermediate DC-DC converters. Each rectifier supplies several DC-DC
converters located in the periphery of the detector or in the counting room.
Switching regulators convert the 48V to appropriated low voltages that are locally42

1.8. PS distribution system of CMS detector

distributed to the detector read-outs. Locating the DC-DC converters in the hostileenvironment near the detector is a disadvantage due to the presence of magnetic
field and radiation [28][29], so custom power supplies are required. Additionally,
the AC/DC units located at the periphery of the detector have to operate undermagnetic field and neutron radiation. The transformer has to operate undermagnetic field and to avoid its saturation, it has to be over-designed. Special
attention should be paid to evaluate the quality of the magnetic material, themagnitude of the external field that the transformer can tolerate without reaching
the saturation region during operation and the impact in the system of thetransformer’s inrush current during start-up.
In general, the subsystems generate the DC High Voltage (HV) using as primarysystem either the 400Hz AC distribution or the 48-V DC. High Voltage power
supplies are, in general, located in remote areas away from the detector due to highenergy particle radiation can induce on these systems catastrophic failures. Long
cables (80-120 meters) will distribute HV to the sensor devices at the front-end
electronics.
Modularity of power supplies (high and low voltage) is the primary requirement. It
will facilitate replacement of damaged units during short-period scheduled accessto the cavern providing a reduction in the time that a part of the sub-detector isdown. The LV cables between the converters and the FEE units are designed to
avoid large voltage drop across them, which means the cross-section is overdesigned for such a current. These lines do not need current protection
supplemental to that built-in into the switching power supplies. Under any
plausible fault condition, the maximum current that the power supply delivers islower than the rated current capability of the wires. The voltage drop estimated for
those conductors is about 0.8/1V. Custom DC-DC converters are designed tooperate under neutron radiation and magnetic fields. All the magnetic devices
included in these units are over designed tolerate the external DC magnetic field.
Electronic component, in particular switching power devices, are selected tooperate reliably under the high-energy neutron radiation. Switching converters arenoisier than linear power supplies. Specifications for ripple and EMC will be very
tight because of the low noise required by front-end electronics.
As summary, the power distribution for the CMS detector has two distinctive
regions: the 400Hz distribution and the DC distribution for the front-end
electronics. The 400Hz distribution area is a conventional AC three-phase
distribution. The appropriated design can be done following the standards andnorms regulating this type of systems. The DC area is characterized by systems withextra low voltages-high current and high voltage (600V-10KV)-very low currents.
Special attention must be put in its design regarding to grounding and safety.

43

1. CMS detector

1.9

Summary

The CMS calorimeter is a very exotic electro-technical environment. Millions of
low-cost measurement channels process very low-level detector signals within aharsh environment (high magnetic field and particle radiation). CMS detector
identifies particle interactions and measures their energy using sensitive detection
devices such as silicon and pixel detectors, optical devices, wire chambers, etc. The
electrical signal detected by the sensors is amplified and processed by the front-end
electronics (FEE), which communicates, via optical link, with the acquisition system
located 120 meters away from the detector.
The detector is organized in five different sub-systems that are located in different
layers. Each of them has a different objective and different signal and power levels.
Also, they are independently powered. These sub-systems are Tracker, ECAL,
HCAL, Muon chambers and super conducting solenoid. The FEE is the mostsensitive part of the detector. It is designed to detect and process signals in the range
of a few fC and digitize them at a rate of 40 MHz. The FEE is characterized by alarge dynamic range. In general, after digitizing the signal, the electronics read-out
serializes it and via an optical link transmits all the information to the acquisition
system.
The front-end electronics into the detector can be considered as an isolated systemwith only a galvanic connection to the exterior through the low voltage and highvoltage power supply system. The CMS detector is primary powered by a 400 Hz
distribution system. These network supplies 3-phase power to the counting room orto the periphery of the detector. At that location AC-DC converters transform the
AC power to DC power required by on LV voltage regulators or intermediate PS
units. These PS units supply the final voltage required by the FEE and detector towork. The CMS detector needs around 6500 power supplies units in order to work.

1.10 References[1] Evans, L.R, "LHC status and plans", Proc. IEEE Particle Accelerator Conference,
19971997, Vol 1, pp 61 - 65, May 1997.
[2] CMS collaboration, "CMS technical proposal", (CERN / LHCC 94-38 LHCC / P1,
1994).
[3] ATLAS collaboration, "ATLAS technical proposal", (CERN / LHCC 94-43 LHCC /P2, 1994).
[4] LHCb collaboration, "LHCb technical proposal", (CERN / LHCC 98 004 LHCC /P4, 1998).

441.10. References[5] ALICE collaboration, "ALICE technical proposal", (CERN / LHCC 95-71 LHCC /P3, 1995)[6] CMS collaboration, "The Tracker project-Technical design report", (CERN / LHCC
98-06, 1998)[7] Gigi Rolandi, "LHCC Annual Review CMS Tracker", Slides for Annual review
meeting, October 2002.
[8] M. Raymond, M. French, J. Fulcher, G. Hall, L. Jones, K. Kloukinas, L-K Lim, G.
Marseguerra, P. Moreira, Q. Morrissey, A. Neviani, E. Noah, "The APV25 0.25 µm
CMOS readout chip for the CMS Tracker", Proc. IEEE Nuclear Science Conference,
Lyon, France, pp 9/113 - 9/118, October 2000.
[9] J. Troska, G. Cervelli, F. Faccio, K. Gill, R. Grabit, A.M. Sandvik, F. Vasey, A.
Zanet, "Optical readout and control systems for the CMS Tracker", Proc. IEEE Nuclear
Science Symposium Conference Record 2002, Vol 1, pp 233 - 237, November 2002.
[10] Giuliano Parrini, " PS Tracker status", Slides for CMS electronic week, May 2003[11] R. D’Alessandro, G. Parrini, "General guidelines for the tracker power supply
system", Private communication.
[12] R. Hammarstrom, "Mutual wire inductance in MSC prototype", Private
communication.
[13] CMS collaboration, "The Electromagnetic Calorimeter project-Technical design
report", (CERN / LHCC 97-33, 1997).
[14] F. Nessi-Tedaldi, "The CMS ECAL project. Overview and status report", IEEE
Transactions on Nuclear Science, Vol 46, N- 3, pp 410 - 413, June 1999.
[15] J. Nash, "ECAL Electronics - System review and overview", Slides for the ESR,
October 2003.
[16] P.Sharp, B. Iliev, I. Nanov, S. Stoenchev, S. Dhawan, B. Betev, W. Lustermann
and M. Pestalozzi, "CMS ECAL Low Voltage", Proc. 9th Workshop on Electronics for
LHC experiments, LECC 2003, Amsterdam, pp 353-357, October 2003.
[17] Paul Aspell, "Conception et mise au point de l’electronique frontale du detecteur depied de gerbe de l’experience CMS" , Thesis, July 2001.
[18] CMS collaboration, "The Muon project-Technical design report", (CERN / LHCC
97-32, 1997).
[19] CMS collaboration, "The Magnet project-Technical design report", (CERN / LHCC
97-10, 1997).

45

1. CMS detector[20] A. Herve, "The CMS detector magnet", IEEE Transactions on Applied
Superconductivity, Vol 10, N- 1, pp 389 - 394, March 2000.
[21] CMS collaboration, "The Hadron Calorimeter project-Technical design report"(CERN / LHCC 97-33, 1997).
[22] T. Shaw et. al., "Front End Readout Electronics for the CMS Hadron Calorimeter",Proc. IEEE Nuclear Science Conference 2002, Viginia, USA, Vol 1, pp 194-197,
November 2002.
[23] T. Shaw et. al., "Electro-optical interfaces for CMS hadron calorimetry", Proc. IEEE
Nuclear Science Symposium 1998, Vol 1, pp 514 - 519, November 1998.
[24] R.J. Yarema, G.W. Foster, J. Hoff, M. Sarraj, T. Zimmerman, "A fast, wide range
charge integrator and encoder ASIC for photomultiplier tubes" Proc. IEEE Nuclear
Science Symposium and Medical Imaging Conference1992, Vol 1, pp 381-383
October 1992.
[25] P. Moreira et. al., "A radiation tolerant gigabit serializer for LHC data transmission",
Proc. 7th Workshop on Electronics for LHC experiments, LECC 2001, Stockholm,
Sweden, pp 145-149, September 2001.
[26] C. Rivetta et. al., "Design considerations of low voltage DC power distribution for
CMS sub-detectors", Proc. 6th Workshop on Electronics for LHC experiments, LECC
2000, Cracow, Poland , pp 388-392, September 2000.
[27] F. Arteche, C. Rivetta, F. Szoncso, "A common 400 Hz power supply distribution
system for CMS Front-end electronics", Proc. 8th Workshop on Electronics for LHC
experiments, LECC 2002 , Colmar, France, pp 203-208 September 2002.
[28] F.Faccio, "Radiation effects in the
http://cmsdoc.cern.ch /~faccio/tutorial.pdf.

electronics

for

CMS-Turorial",[29] C. Rivetta, B. Allongue, G. Berger, F. Faccio, W. Hajdas, "Single event burnout in
DC-DC converters for the LHC experiments", Proc. 6th European Conference on Single
event burnout in DC-DC converters for the LHC experiments, pp 315-322
September 2001.

46

Chapter 2
Grounding & Shielding

Safety, system protection and performance are the three main reasons to earth asystem. Not all electronic equipment needs to be connected to earth to work,
satellites are an example. Sometimes wrong grounding configurations, oriented tosatisfy the special power and performance requirements of electronic loading
equipment, can compromise safety rules generating dangerous situations for
personnel and equipment. Personnel safety, equipment safety and performance
grounding issues have to be analyzed together. In any case, safety rules must not be
violated. All these concepts as well as the relations between ground issues need to
be clarified before passing on to a higher level in the control of the electromagnetic
emissions of the system. Along the present chapter, concepts like safety grounding,signal or ground reference plane as well as grounding topologies to achieve electricsafety and good performance of the system, are presented. The characteristics ofshield structures, shielded power cables as well as their connections to ground arealso presented. Finally in the last section, some recommendations to design the
CMS detector grounding are included. The grounding design of the detector
constitutes the first of several steps to be followed for the detector integration undera electromagnetic compatibility plan.

2.1 Grounding generalities
As described in [1] [2] and [3], proper grounding techniques are necessary for safety,
equipment operation and performance. The integrity of both the facility grounding
and the proper equipment operation depends on the proper bonding of thegrounding electrode system, proper system grounding of service equipment andseparately derived sources and proper equipment grounding for operationalfrequencies (DC or AC-mains) as well as higher frequencies. It is recommended thatthe grounding design and installation be compliant to all applicable codes andstandards [4][5]. The grounding is not designed as an active component of thepower supply (PS) distribution system, hence this path must be free of any
operational current.

47

2. Grounding & Shielding

Metal parts of equipment enclosures, racks, raceways and equipment grounding
conductors susceptible of being energized by electrical currents (due to circuit
faults, electrostatic discharge, and lightning), must be effectively grounded forreasons of personnel safety, fire hazard reduction, equipment protection andequipment performance [6][7]. Grounding these metallic objects facilitates theoperation of over-current protective devices during ground faults and permits
return current from electromagnetic interference filters and surge protective
devices, connecting line to ground or line to chassis, to flow in proper fashion. All
metallic conduits and raceways in areas containing electronic load equipment haveto be carefully bonded to form an electrically continuous conductor.
All mechanical equipment in the electronic equipment areas should be carefully
bonded for an electrical safety and for noise current control. Such equipment should
be grounded or bonded to local building steel using direct or higher frequency
grounding and bounding means. When located in the same area as the electronicload equipment, mechanical equipment should be bonded at multiple points to thesame ground references as the electronic load equipment. Heating, ventilation, airconditioning, process cooling equipment, related metal piping and electricalconduits are recommended to be bonded to the same ground reference serving theelectronic load equipment.
Once the grounding complies with the safety rules, the ground connections have tobe improved to obtain a good performance of the system. It is important to keep inmind that for operational reasons one should imagine that at low frequencies theground system is a kind of low resistive divider in which all noise currents usuallyflows everywhere. However at high frequency the ground impedances start toincrease mainly due to the inductive effect, flowing the noise only through thelowest impedance path. Therefore, the ground connections should be tackled in two
steps. The first step is oriented to low frequency currents where it is important toavoid ground loops because once these currents are established they can flow
everywhere, decreasing the performance of the system. The second step is focusedon the noise at high frequency, where one could imagine that everything in thesystem is connected (through real or parasitic impedances) designing the ground
path and connections in an specific way that gives a low impedance path to HF
noise currents and thus avoiding the flow of these currents through sensitive partsof the system.

48

2.2. Grounding for fault and personnel protection.

2.2 Grounding for fault and personnel
protection.
2.2.1 System grounding or earthing
System grounding or earthing involves the ground connection of power services
and separately derived systems. They include generator, transformers,
uninterruptible power systems (UPS). The system earthing is the intentional
connection of a circuit conductor (typically the neutral on a three phase, four wire
system - Protective earth (PE)) to earth. The purpose of the system grounding [3] isfor electrical safety of personnel and equipment as well as fire safety reasons. Safety
is basically governed by the electrical codes and standards as adopted bygovernment agencies and commercial entities. System grounding also impacts theperformance of electronic load equipment for reasons related to the control of thecommon-mode noise and fault currents, however the personnel and the equipment
protection is the primary task of the grounding as it is described in [8].
The grounding of power systems is, from a safety standpoint, oriented to limit thepotential difference between grounded objects, to provide a good operation ofover-current protective devices in case of ground fault, to stabilize the phasevoltages with reference to ground and to limit transient voltages due to lightningand load switching.
There are two basic requirements for grounding power services and separately
derived systems or sources (transformer, generators, UPSs, etc.). The firstrequirement is to bond the neutral or secondary grounded circuit conductor to theequipment grounding terminal or bus. For power services entrances, the incoming
neutral conductor is connected to the equipment grounds bus in the switchboard bymeans of the main bonding jumper. For separately derived sources, the neutral
must be bonded to the equipment grounding terminal or bus. The second
requirement is that the equipment grounding terminal or bus must be connected to
the nearest effective grounded electrode by means of the grounding electrodeconductor. To illustrate the grounding connection of a separately derived source,
figure 2.1 shows the grounding connection for an isolation transformer. If noeffective grounded electrode or building steel is available, then the separately
derived source should be connected to the service entrance grounding point via agrounding electrode conductor installed in the most direct and shortest path
practicable. In the case that metal interior piping is present near the separately
derived source, a supplemental grounding electrode conductor should also be
installed from the equipment grounding terminal or bus of the separately derived
source to the metal interior water piping.

49

2. Grounding & Shielding

Figure 2.1: Isolation transformer grounding layout.
From a performance standpoint, solidly grounded power systems arerecommended practice to ensure the existence of effective conductive paths for the
return current of filters and surge protective devices connected line to ground orline to chassis. These filters and surge protective devices may be an integral part ofthe electronic load equipment or may be separately mounted devices located in thebuilding electrical distribution system. It is recommended in the design to aim at
the lowest reasonable impedance between the load equipment containing a filter orsurge protective device and the associated power system source. Low-inductancewiring methods should also be used.

2.2.2 Equipment grounding
As described in [9][10][11] and [12], electrical or electronic circuits do not need to beearthed in order to work. Satellites, spacecraft and mobil phones, all work properly
without earthing. The earthing or system grounding is a requirement necessary forsafety. Electrical safety concerns all electrical design work. Safety requirements
cannot be compromised to satisfy the special power and grounding requirements ofelectronic loading equipment. One should always try "to make the system design safe
and then try to make it work". This ground philosophy is widely explained in [7], [13]and [14].
The term "equipment grounding" refers to the connection to power system groundof all non-current carrying metallic parts of a power system that may come into
accidental contact with circuit phase and neutral conductors. These metallic parts
include raceways, conduits, equipment grounding conductors, equipment50

2.2. Grounding for fault and personnel protection.

enclosure and racks. All these items are ultimately grounded together at the
grounding electrode of the power service or a separately derived system.
Equipment grounding is required for both personnel safety and power systems
protection. From a personnel safety point of view, properly grounded systemcomponents minimize potential differences that may exist between various systemcomponents under transient and fault conditions. From a system protection
standpoint, properly grounded system components provide a low impedance pathfor ground fault currents and promote the timely operation of over-current
protective devices in case of ground faults.
There are two different grounding configurations associated with the connections of
the equipment to the PE point; the standard ground configuration and the insulated
ground configuration.
• For the standard equipment grounding configuration, a supplementary ground
conductor provides an additional low impedance ground path in parallel withthe metallic conduit or raceway from the equipment to the power system.
• For the isolated grounded configuration, the electronic or electrical load is notconnected to the metallic enclosure and this metallic enclosure is earthed
through an independent conductor to the start point of the safety ground. The
insulated equipment grounding conductors run with the other circuit
conductors feeding electronic load equipment. In an isolated groundconfiguration, the additional equipment grounding conductor provides the solegrounding path from the electronic or electrical load equipment to the powersystem or separated derived system.
In either case, the insulated equipment grounding conductor should run in thesame raceway or conduit as the phase and neutral conductors. Grounding
configurations provide equalization of potentials between grounded objects at theoperative frequency. As the frequency increases, other grounding means must be
considered to cover the high frequency range.
Robust design of the electronic load equipment for immunity to disturbances on the
grounding circuit is a good method to get a good grounding [15]. Particularly for
distributed computing and telecommunication electronic loads, using opticalsignaling interfaces reduces the sensitivity to disturbances on the ground circuit. As
it is implemented in the CMS detector, the processed data from the detector istransmitted out of the detector via optical devices, and the slow control signals are
isolated from the detector via opto couplers.
2.2.2.1 Standard equipment grounding
The standard equipment ground configuration [13][14][16] uses one equipment
grounding conductor (PE or G), in green color, running with the phase and neutral51

2. Grounding & Shielding

conductors to supplement grounded
configuration is shown in figure 2.2.

metal

raceway

andconduit.

This

Figure 2.2: Standard ground configuration.
The standard equipment grounding conductors are usually sized according to thetable of standards for equipment grounding conductors and are properly connected
and bonded to each metal enclosure that it passes through from the separately
derived system or power service to the electronic load equipment. These metal
enclosures include all distribution panel boards, safety switches, circuit breakerenclosures, transformers and branch circuit panel boards, as well as pull boxes,
junction boxes and metal outlet boxes.
The conduit and raceway system may depend on the integrity of mechanicalconnections at conduit and raceways joints, panel boards, junction boxes and at thereceptacles themselves. The non-uniformity and bad performance of the grounding
paths can compromise personnel safety as well as the operation of surge
suppressors and filters located in the electronic load equipment. In addition,
currents flowing on grounded surfaces may take less desirable paths, such us
through load equipment and associated data cables. The purpose of the installed
equipment grounding conductor is to increase the reliability of the grounded metal
conduit system.
2.2.2.2 Isolated grounding
The isolated grounding configuration [16][17] uses an insulated equipment
grounding conductor, typically green color with yellow stripe, running with thephase, neutral and PE conductors from the electronic load equipment to theequipment grounding terminal of the power system or separately derived system.
As opposed to the standard equipment grounding configuration, this additional
insulated equipment grounding conductor typically connects the isolated ground522.2. Grounding for fault and personnel protection.

receptacle (IGR) only to the equipment grounding terminal or bus of the powersystem source or separately derived system. This equipment grounding conductor
extends radially downstream to the chassis of the electronic load equipment
without contacting any grounded metal surfaces such as metal conduits and race
ways, panel boards, and outlet boxes for receptacles. This configuration is shown infigure 2.3.

Figure 2.3: Isolated ground configuration.
The isolated equipment grounding conductors are sized according the table ofstandards for equipment grounding conductors and are properly connected andbonded to the metal enclosure at load level as explained in the standard equipment
grounding. This type of equipment grounding configuration is only intended to be
used for reducing the common mode (CM) electrical noise on the electronic loadequipment circuit as described in [13] and [17].
This type of configuration may produce better or worse noise conditions than whena standard equipment grounding configuration is used to serve electronic loadequipment. Noise effects will be somewhat proportional to the overall length of thecircuit [17]. Under power system fault conditions, the potential difference between
the electronic load equipment and grounded objects may be sufficient to cause asafety hazard or to disrupt the electronic load equipment performance.
The isolated grounded configuration is only directly applicable to metal-enclosed
wiring means and has no useful purpose with nonmetallic wiring systems.
Non-metallic wiring systems are at least partially constructed as if they are isolated
grounding types, since no metal conduit or raceway is involved in the wiring pathor is bonded to the equipment grounding conductor of the circuit. In any case, thenon-metallic wiring system does not provide electromagnetic shielding for theenclosed circuit conductors and should not be used.

53

2. Grounding & Shielding

The application of the isolated ground configuration may provide beneficial effects
to circuits supplying electronic load equipment that do not otherwise connect togrounded objects. However, if the electronic equipment contains other connections
to grounded objects as in the standard ground configuration, the performance of theisolated grounding configuration decreases. These connections to ground may beeither intentional or unintentional. Typical examples of these connections areinterconnections of various equipment through grounded shields of data cables andbonding of equipment chassis to grounded metal equipment racks.

2.3 High frequency grounding configuration
The grounding configurations described above provides the necessary connections
to ensure the overall electrical safety of the system. When two or more components
of an interconnected system are installed in an area where there is a physical space
between them and across their separation data input/output cables and inter-unit
power circuit cables (DC, AC or both) are routed, there exist indirect bounding
problems ranging from DC to several tens of MHz or higher that can compromise
the system performance. A reasonable grounding system has to be designed,
without compromising electrical safety, by defining a ground reference structure
over a broad range of frequencies. For separated equipment or units is necessary toplace all of them on a single sheet of metal in the form of a signal reference plane
and then to use direct grounding or bonding techniques to connect the entire
perimeter of the base of the unit to signal reference structures (SRS). These reference
structures can be built as reference planes (RP) or grid structures (GS). Linear
brazing or welding around the perimeter of the unit’s base is one method forgrounding to the structures. However, such direct connections are not often
practical, and the next best approach is to use multiple indirect bonding straps ofminimized length to connect individually the plane to each unit locally.
The SRS is not intended to be dielectrically or galvanically insulated or isolated
from the safety grounding conductor (PE) system that is part of the fault/personnel
protection grounding system. The principal purposes of the SRS are:
• To enhance the reliability of signal transfer between interconnected items ofequipment by reducing inter-unit common electrical noise over a broad band of
frequency.
• To prevent damage to inter-unit signal circuits by providing a low-inductance,and hence, effective ground reference for all of the externally installed AC andDC power, telecommunications, or other signal level, line to ground/chassis
connected equipment that may be used with the associated equipment.
• To prevent or minimize damage to inter-unit signal circuits and equipment
power supplies when a power system ground fault event occurs.

542.3. High frequency grounding configuration

The need of an SRS is minimal when all of the inter unit signal andtelecommunication circuits are interfaced to the associated electronic equipment via
optically or isolations transformer coupled means. It is used in CMS detector, wherethese interfaces have good common mode voltage breakdown characteristics.
However, the need for an SRS may easily rise to that of a requirement in the event
any of the following three conditions are given:• When the logic AC-DC power supplies used in the associated electronicequipment are installed with one of the terminals connected to the equipment’s
metal frame/enclosure. This is typical and recommended practice in theequipment industries.
• When the signal-level circuits and logic AC-DC power supply common
terminals are dielectrically insulated or galvanically isolated from the equipment
ground against recommended practice, and are instead connected to an
insulated ground terminal that is intended for connection to an externallyinstalled signal ground reference circuit.
• When there are actual performance problems occurring with the equipment,
which can be assigned to common mode electrical noise or similar common
mode interference related to the equipment’s existing grounding system,
whatever its design, or the signal-level
Improved HF grounding for data signaling cables between (non-contiguous) areas
can typically be accomplished by reducing the open loop area enclosed by the cableand its grounded surroundings. This is typically accomplished via the use of metal
conduit or electrically continuous, solid-bottom, metal cable tray, wire-way, or
similar forms of signal transport ground-plane constructions.
The main advantages of the SRS are:• Low-impedance return path for RF noise currents• Containment of EM (noise) fields between their source (cable, etc.,) and theplane• Increased filtering effectiveness of contained EM fields• Shielding of adjacent equipment
An SRS may be typically constructed using one the following methods (indecreasing order of effectiveness), which are well described in [16]:• Solid covering of sheet metal (RP).
• Grid of cooper straps (GS).
• Grid of cooper and aluminium wire (GS)• Rised flooring substructure (GS).
Hybrid forms of SRS employing mixtures of signal reference grid and signal
reference planes for varied construction and improved overall performance are also55

2. Grounding & Shielding

useful. They are used where the benefits of each type of SRS are needed for thecollective support of a variety of interconnected electronic load equipment that issusceptible to CM noise current.
The use of RP may be recommended for some applications where the subject
system operates at a higher frequency than the typical GS signal design cut-off
frequency. Although the RP does not offer a zero impedance path across its surface,it does offer several orders of magnitude of improvement over any signal or groundof large-cross section wires that might be used for bonding between units. The
general equation for determining the impedance between two points on a plane is:2⋅π⋅d
Z gp = ( R DC + j ⋅ Z RF ) ⋅ 1 + tan -----------------λ

(2.1)
2

where R DC is the DC resistance of the ground plane using the Ω ⁄ m , Z RF is the2

impedance of the ground plane in Ω ⁄ m . and d the distance between two points.
Accordingly, the impedance between two points on a ground plane should beapproximately equal to the ohms per square value of the material of which it iscomposed and of the length such that the distance d is short compared with wave
length λ of the highest frequency of interest in the design. Finally, the groundplane must be at least as wide as d. Based on this concept, it may be seen that the
impedance value between points across dimensions d is constant at DC level andgradually rises as the applied frequency of the current is risen. Finally, isotropic risein impedance may be expected at the first resonant point encountering 0.25λ on theplane and at each succeeding odd-order multiples 0.5λ of the first resonant point
generally approaching the expected impedance along the projection of Z N . This isshown in figure 2.4. The most favorable area of operation for the ground plane is in
the range where 1 < 1 ⁄ 20 ⋅ λ at the highest frequency of concern.

Figure 2.4: Frequency response of a ground plane.

562.3. High frequency grounding configuration

Often the solid form of a ground plane cannot be obtained and a mesh or grid form
must be a substitute for it. GS may be thought of as a plane in which holes, whose
dimensions have been kept below a critical value, have been placed in a repeating
pattern. In general only grounding grids using squared cells are recommended foruse for HF currents. One of the most important issues related to grounding grids
operating in the HF area, as opposed to DC and AC power frequency areas, is theneed for minimizing the reactance in the path of the grounding grid instead of theresistance.

2.3.1 Multipoint and single point grounding connections
There exist different grounding configurations according to the connection of thedifferent components or units to a signal reference system. Although they are well
explained in [14] and [16], some details of the most important configurations, thesingle point and the multipoint connections are now presented. The determination
to use single-point grounding or multipoint grounding typically depends on thefrequency range of interest. Analog circuits with signal frequencies up to 300 kHz
may be candidates for single-point grounding. Analogue/Digital circuits withsignal frequencies in the MHz range should use multipoint grounding.
2.3.1.1 Single point
As it is shown in figure 2.5, the single point grounding is the easiest configuration ofgrounding. It is very useful in small systems. Nevertheless, if the real configurationof the single point grounding is analyzed (figure 2.6), it is the most undesirable
ground system from the EMC point of view. Firstly, because of the big inductancethat the ground path presents, the ground loops and the stray capacitanceassociated with the system. Secondly, because of the unreasonable large amount ofwire necessary.

Figure 2.5: Single point grounding configuration.

57

2. Grounding & Shielding

Figure 2.6: Real single point ground configuration.
Single point grounding cannot be easily implemented in SRS since these structures
depend on a multiplicity of connections. Single-point grounding is usually
implemented with a physical bus or bulkhead form of construction, where all
conductors are connected prior to entering or leaving the signal reference grid area.
A potential violation is to have a single-point ground area where one additional
grounding connection occurs at a remote point within the electronic loadequipment that is normally designed to be grounded only at one point. Such aconnection may be intentional or unintentional. This configuration would provide awell defined and concentrated current path through the electronic load equipment,
which could cause performance problems or component damage. More details
about this kind of designs are shown in [19].
2.3.1.2 Multipoint
The recommended practice for signal reference systems (SRS) is multipoint
grounding. Multipoint grounding requires that all metallic objects crossing orintersecting the signal reference system are effectively bonded to it. This type ofgrounding is shown in figure 2.7

58

2.3. High frequency grounding configuration

Figure 2.7: Multi point ground configuration.
Multipoint grounding to the SRS minimizes the possibility for all types of electricalcurrents flowing in the signal reference grid to be unwanted onto a few conductorsof the signal reference system (this controls near-field conditions and potential
difference as well). Based on this type of grounding, it is strongly recommended toseparate at the origin the ground (reference or return) of analog signals, digital
signals, power supplies and racks and afterwards connect them in a specific waythat minimizes the differential of potential between different grounds. This is valid
at system or board level. More details can be found in [16] and [18].

2.3.2 Bonding and straps
All equipment, especially electronic load equipment, should be connected to thesignal reference system through a low-inductance path. The connections can bemade at both the facility and equipment level by means of indirect bonds or directbonds. The indirect bond is the connection in that a wire, strap or bus bar is used tointerconnect the items or units together, whereas in the direct case, the bonding
connection is made by direct connection using screws or solder joints. Indirectbonding is discussed in this section.
The purpose of bonding two items is to equalize potential between the items and toensure that a minimum of potential can be developed between them over a broadfrequency range, typically from DC to several tens of MHz and beyond. To achieve
this goal, the design of bonding strap must be one providing low impedance acrossthe desired frequency range for AC and very low resistance for DC. There is always

59

2. Grounding & Shielding

resistance in the metallic path and for high frequency currents the impedance
increases due to skin effect and inductive effects.
Bonding path installed on the site and intended for use at or near the power mainfrequency, such as for ground fault currents, need not consider much more than theresistance of the material being used together with its heating and fusing
characteristics. Thus simple wire resistance and capacity tables are normallyadequate for the determination of the path’s performance with regard to the sizing
and minimizing of ground fault voltage drop across the bonding path. However for
longer ground fault path lengths, wire impedance tables that account for theslightly increased impedance due to skin effect and reactance at the power
fundamental frequency must be used. When unwanted AC signals are present in
the grounding / bonding path, things are somewhat different. In these cases theneed to minimize the potential developed across the bonding path under highfrequency (HF) voltage and current conditions of all kinds is of major importance.
This need entails the consideration of the path impedances and with the reactanceof the bonding path being of greatest importance.
In general the total impedance of a bonding conductor is given by:
Z = R DC – AC + j ⋅ ω ⋅ L(2.2)

The most important term of the above equation is the inductance L. The magnitudeof the inductance L is calculated based on the concept of ’partial inductance’. The
partial inductance assumes that a current flowing through a segment of unitarylength defines a virtual loop, where the opposite segment is at infinity. Based on
figure 2.8, the partial inductance L pij is the ratio of the magnetic flux penetrating
the surface between segment i and the infinity and the current j that produces it.

Figure 2.8: Partial self and mutual inductance loops.
This definition is used to represent the eventual loop interaction without the priorknowledge of the current loop. Based on this definition it is possible to calculate
partial self and mutual inductance. The total or loop inductance is equivalent to thesum of the partial self-inductance and mutual inductances of the segments, asshows equation 2.3.

60

2.3. High frequency grounding configurationNL =∑ ±Lpij

(2.3)

j=1

where the loop contains a total of N segments in which the loop has been divided.
Each segment supports a current I j , and the sign of each term corresponds to therelative orientation of the currents assigned to segments j. A detailed definition andanalysis about partial and total inductance can be found in [12][20][21][22] and [23].
Based in the definition above mentioned, it is possible to calculate the partial
inductance of a single round bounding.
4⋅l
L µH = 0.002 ⋅ l ⋅ ⎛ ln ⎛ ---------⎞ – 0.75⎞⎝ ⎝ d ⎠⎠

(2.4)

where l is the length in centimeters and d is the diameter of the wire also in
centimeters. Basically the value of this inductance depends on the length and thediameter of the bounding, increasing the diameter of round conductors has alimited effect on the HF impedance presented by the strap. On the other hand, if a
rectangular bonding strap is considered as a flat-strap, the following equation
applies:b+c2⋅l
L µH = 0.002 ⋅ l ⋅ ⎛ ⎛ ln ⎛ ------------⎞ ⎞ + 0.5 + 0.2235 ⋅ ------------⎞⎝ ⎝ ⎝ b + c⎠ ⎠
l ⎠

(2.5)

where l is the length in centimeters, b is the width of the wire in centimeters and c isthe thickness of the strap. The length to width ratio of a strap may be significantly
altered to benefit a bonding path’s performance, whereas altering the diameter of around wire cannot produce useful improvements. Basically, it is much better to do
the ground connections with flat-straps than with round conductors. It is also wellknown that the inductance increases by winding a conductor into a coil or similar
form where the magnetic lines of force along the conductor may be concentrated
causing an interaction among them. Thus, it must be expected that the best
performance is achieved on a bonding path or with only a minimum and gentle
bends in the winding conductor. Loops and sharp angles must be avoided unless an
inductance is deliberately being created at the point of path discontinuity.
Bounding straps may self-resonate or may resonate with the stray or parasitic
capacitance between the equipment and ground. The bonding strap exhibits highimpedance near resonance and at resonance it looks much like an open circuit.
Hence operation on this area must be avoided. The variation of the impedance withthe frequency and the equivalent circuit of the strap are shown in figure 2.9.

61

2. Grounding & Shielding

Figure 2.9: Strap’s equivalent circuit and frequency response impedance.
Radiation to or from a bonding strap is not to be overlooked as a possible problem.
However this is a problem only at and very near the actual point of parallel
resonance in the strap. Typically any radiator, that is kept to 1 < 1 ⁄ 20 ⋅ λ , will notact as a good radiator or receptor. As expected this consideration can also pose
major design limitations on the overall length of bonding straps or any conductorused as a ground connection.
Minimizing the impedance between the items being bonded is always important.
The use of a single bonding strap between items may make this difficult toaccomplish. If multiple, parallel straps are used, the impedance may be reduced
between items according to:1
L T = --------------------------------------------------------1 ⁄ L1 + 1 ⁄ L2 … + 1 ⁄ Ln

(2.6)

Resistance (AC and DC) and inductance are affected in much the same way.
However in the case of the inductance, unless the mutual inductance is also kept toa minimum between parallel straps, they will not act as independent inductances inparallel, the mutual inductance in that case must be considered also. Spacing must
therefore be maintained between straps for at least the length of the longest strap
employed if this problem wants to be avoided. A detailed analysis of this concept isshown in [16]. Grounding straps should be as short as possible to minimize
inductive reactance in the path. The use of at least two bonds widely spaced aparton the same item of equipment is recommended to further reduce the reactance ofthe grounding path. These straps should be of different length such that they will
have different self-resonant frequencies. The straps should never be folded orcoiled, nor bent into curves. Even in equipment, lineups where the equipment is62

2.4. Shielding

bonded together, the recommended practice is to bond each enclosure to the signal
reference grid with its own strap or two if it becomes easier. Flat foil strips, which
are relatively wide in relation to the length, are the recommended practice.
Connections to the equipment frame or a supplied grounding terminal are critical.
Paint or other surface contact inhibitors should be removed before bonding straps
are directly attached to metal enclosures or cabinet surfaces. Subsequently, theconnections should be properly treated to inhibit rust, corrosion and moisture.

2.3.3 Signal reference structure for non-continuous area
The signal reference grid or signal reference plane is appropriate for a single twodimensional area and nearby contiguous areas, but is impractical and not as
effective between widely separated areas or buildings, as it is the case of the CMS
complex. Recommended practice is to augment the circuits with surge protective
devices. Other methods (e.g. optical isolators or suitable wide band common modecurrent filters) can also provide increased noise and surge immunity for theinterconnected telecommunication, data, and signal circuits.

2.4 Shielding
2.4.1 Introduction
The objective of electromagnetic, electric and magnetic shielding is to provide asignificant reduction or elimination of incident fields that can affect sensitive
circuits as well as to prevent the emission of components of the system fromradiating outside the boundaries limited by the shield. The basic approach is tointerpose between the field source and the circuit a barrier of conducting or
magnetic material.
Shielding effectiveness can be defined as the reduction in magnetic, electric orelectromagnetic field magnitude caused by the shield. The effectiveness of a shielddepends on the shield material as well as the characteristics of the incident field (faror near field), which is defined by the distance between the source and the victim.
So, it is found that techniques for shielding depend on the type of source; whether
the source is a magnetic field, electric field or electromagnetic field source. The
shielding effectiveness (S) in dB, can basically be calculated as the sum of threecomponents, namely, reflection loss (R), absorption loss (A) and a correction factor(B) used in special cases to consider multiple reflections in the shield.
S = A+R+B

(2.7)

63

2. Grounding & Shielding

Each component has a different expression and value depending on whether theincident wave is magnetic, electric or electromagnetic field. Although analyzed indetail in [12][24][25][26] some issues are presented in this section. The presence ofholes and joints can decrease the effectiveness of a shield and its analysis is welldetailed in [12] and [25]. The present study does not provide further analysis on thesubject though it should always be taken into account during the design phase.
Other important point analyzed in this section is the shielding of cables. Special
attention is paid to the connection of the shield at either one or two ends of a cableand the implications that these connections have in the performance of the system.

2.4.2 Electromagnetic field shielding
Although any radiated wave is an electromagnetic wave, the term electromagnetic
wave is generally used to describe a far-field, plane wave, where the ratio between
the electric field and the magnetic field is defined by the characteristic impedance ofthe free space (Zo=377 Ω ). When an electromagnetic wave passes through a
medium, two phenomena, known as absorption and reflection losses, are present.
In the former, induced currents generate ohmic loss, heating the material, andproducing an exponential attenuation of the amplitude in the direction of the wave
propagation [24]. In the latter, when a field arrives at an interface between two
media, part of the field can be reflected, introducing new losses. Basically the total
loss is a combination of these two losses. The multiple reflection correction term isnormally not considered for this type of waves as the reflection loss is high and thecorrection term is small.
Assuming the electromagnetic wave propagates perpendicular to the shield surface,
the absorption and reflection losses (in dB) can be calculated as:
A = 131.4 ⋅ t ⋅ f ⋅ µ r ⋅ σ r(2.8)µ r ⋅f
R = 168 – ⎛ 10 ⋅ log -----------⎞
⎝
σr ⎠

(2.9)

where t is the thickness of the shield in m., f the frequency, µ r the relative
permeability and σ r the conductivity. This equation is in close relation to the skindepth of the material. The reflection losses decrease with the frequency, whereas theabsorption losses increase due to the skin effect. Based on these equations, it ispossible to state that the reflection loss is the primary contributor to the shielding
effectiveness at low frequencies. However, at higher frequencies the absorption loss
is the primary contributor to the shielding effectiveness at high frequencies.

642.4. Shielding

2.4.3 Electric field shielding
In the near-field, the relation between magnetic field and electric field is notdetermined by its characteristic impedance in free space (377 Ω ). The basic
mechanisms of shielding observed for far-field sources are valid for near-fieldsources, but the type of source is critical for determining the shielding methodology
to apply. For sources dominated by high voltages the predominant near-field ischaracterized by an electrical field, whereas for sources with high currents thedominant near-field is a magnetic field.
Basically, electric shielding consists of conductive barriers, metal enclosures, metal
conduits or cable coverings around circuits. The spatial electric shield acts as acapacitive voltage divider between the field source and the circuit. As it was
defined before, for a field propagating perpendicular to the shield surface, theeffectiveness of the shield against an electric field is defined by the absorption andreflection losses, in which the absorption term is unaffected by the source, being
equal to the equation 2.8. However, the reflection loss factor depends on the source
and in the case of an electric field can be approximated by:
σr ⎞
⎛
-⎟
R = 322 – ⎜ 10 ⋅ log ---------------------3 2⎝µr ⋅ f ⋅ r ⎠(2.10)

where r is the distance between the source and the shield. When the distance isundetermined, the equation showed for the far-field reflection could be usedinstead, as electric near-field reflection losses are always lower than, or equal to, thefar-field reflection losses, hence introducing a certain safety margin. Based on thoseequations, conclusions for fields with dominant electric component are very similar
to those attained for electromagnetic fields. The reflection loss is predominant at
low frequencies, while absortion loss is predominant at higher frequencies.

2.4.4 Magnetic field shielding
In the case the magnetic component of the near-field is dominant, and the wave
propagates perpendicular to the shield surface, the absortion losses are the same as
those for far-field, defined by equation 2.8. However, the reflection losses aredifferent and can be approximated as:2

σr ⋅ f ⋅ r ⎞⎛R = 14.57 – ⎜ 10 ⋅ log ---------------------⎟µr ⎠⎝(2.11)

Based on this equation, the reflection loss decreases for decreasing frequencies, andis lower than the reflection loss for the plane wave reflection. So, reflection losses

65

2. Grounding & Shielding

are usually negligible for lower frequencies and absorption losses are small for low
frequencies too. This fact forces the use of different shielding techniques against low
frequency magnetic fields [27][28][29].
Basically, there exist two different ways for shielding against low frequency
magnetic fields [12][24].
• Deviation of the magnetic flux with high permeability material.
• The shorted tuned method, which consists in the generation of opposing fluxesthat cancel the magnetic field in the area of interest.
To deviate the magnetic flux, it is recommended the use of magnetic material
instead of conductor material because it increases the absorption losses, hence
improving the attenuation of the magnetic field (as it is the primary shielding
mechanism at low frequency against magnetic fields). However, when a magnetic
material is considered as material shield, two properties of this material, which
introduce some limitations, have to be taken account:• The permeability of a magnetic material decreases by increasing the frequency.
• The permeability of a magnetic material decreases by increasing the magnetic
field strength.
The former depends only on the material, the latter depends on the material and thesection of the magnetic circuit. For example mu-metal material has a permeability ofover 10000 from DC up to 1 kHz, however at 20 kHz the permeability is not larger
than cold-rolled steel. The working conditions of the shield is an important point as
it has a serious impact in the selection of the shielding material.
As a summary a magnetic material such as steel or mu-metal makes a better
magnetic field shield at low frequencies than does a good conductor such asaluminium or copper. However at high frequencies, good conductors provide better
magnetic shielding. For non-magnetic material the shield effectiveness increases
with the frequency, therefore, it is recommended to calculate the attenuation for thelowest frequency of interest. On the other hand, for magnetic materials the shieldeffectiveness may reduce due to the decrease of the permeability with the frequency.

2.4.5 Cable shielding
In the present section the main characteristics of a shielded cable as well as theimplications of the shield connections in the performance of the system arepresented. The main goal of a shield is to avoid perturbing fields to penetrate into
the internal conductors or perturbing currents in the central conductors to radiate.
The type of shield material and the shield connections have a direct influence in theperformance of the shield.

662.4. Shielding

A shield is made of non magnetic materials, which as it has been shown in theprevious section, has very bad properties to attenuate magnetic fields, and specially
low frequency magnetic fields. A shield should never be considered as a goodshield against magnetic fields unless a current is allowed to circulate through it,
which can create an opposite field that tends to cancel the perturbing field. These
currents can only circulate if both sides of the shield are connected.
When a shield is not connected to ground or is connected only in one side, theshield attenuates only the electric component of the incident field. In thisconfiguration, the shield can be considered as a capacitive voltage divider as it isshown in figure 2.10. Connecting the shield to ground is required to define thecapacitive voltage divider. The shield side closest to the electronic system should begrounded in this case.

Figure 2.10: Shield connection.
This type of connection has been devised in order to prevent conductive "groundloops" from being established, which would cause unwanted current to flow in ashield grounded at more than one place, e.g at each end. This implementation only
avoids ground loops when dealing with low frequency signals. At higher
frequencies, the parasitic components capacitance between the cable’s shield andthe surrounding areas allows flowing currents through the shield. Additionally, theefficiency of the cable shield working as an antenna (receptor or transmitter) starts
to play an important role in the emission, propagation and reception of the noise in
the system. When a cable’s shield is grounded at one end only the opposing end of
the shield is under grounded and it can represent a fire and shock hazard if thecable’s shield becomes energized. Reasons for being energized could be; AC powersystem ground faults, accidental contact of the shield at some point along its lengthwith a conductor of another system or higher voltage, lightning, etc.
When EMI protection of the signal from near-field with strong magnetic fieldcomponent is required, it is necessary to ground the cable’s shield at both ends. It
allows flowing a shield current induced by the near field magnetic component. This67

2. Grounding & Shielding

shield current generates a magnetic field opposite to the perturbing magnetic field,
attenuating its effect on the central conductors of the shielded cable. This effect does
not have any adverse effects on the cable shield attenuation to electric fieldcomponents.
The attenuation to electric field and magnetic fields is quantified in shielded cables
by two characteristic parameters as the transfer admittance and the transfer
impedance, respectively. The transfer impedance of a shield is defined as the ratio
between the voltage per unit length generated by the circuit formed by the shield
and the conductors inside the shield and the shield current. The transfer impedance
gives the open-circuit voltage developed between the internal conductors and theshield for one ampere of shield current, in a cable of 1 meter long [25]. On the other
hand, the transfer admittance of a shield is defined as the ratio between the current
per unit length in the conductors inside the shield and the voltage between theshields and the external structure. The transfer admittance gives the short-circuitcurrent induced in the internal conductors (when the internal conductors areshorted to shield) for one volt between the shield and the external structure, in a
cable of 1 meter long [25]. Both the surface transfer impedance and the surface
transfer admittance are characteristic parameters of shielded cables and are studied
more in detail in chapter 4.
Cable shields grounded at both ends can carry unwanted shield currents due topotential difference between the two grounded ends of the shield. In general, thesecurrents will be related to the power system’s fundamental and harmonicfrequencies. DC and low frequency currents in the shield, as described above, canbe eliminated or significantly attenuated by placing a blocking device between theshield and its ground connection point at one end. For example, a series-connected,
back-to-back arranged stack of rectifier diodes or capacitors can be used. When highfrequency capacitors are used at one end of the shield, the capacitor blocks the low
frequency currents but allows high frequency current to flow through the shield toincrease the shield attenuation to magnetic fields at high frequency. Back-to-back
diodes or surge diodes can be used when the connection between the shield and theground point has to be established if the potential between these two structures ishigher than the voltage drop of the semiconductor devices [14].
The cable’s shield also offers a return path to common currents flowing through thecentral conductors. The return path is particularly important in shielded power
cables to return the common mode currents generated at the output terminals ofswitching power supplies. If a regular cable without shield is used to connect thesepower supplies to the load, common mode current will establish between the
conductors and the signal reference system (SRS), existing a large radiation loop forthese currents. In order to be effective, shields must be grounded via
low-impedance paths at the frequencies of interest. Long shields need to be
grounded at multiple locations along their length [12][16][17]. To be effective,
shields should be grounded using especial connectors that bonds the total

68

2.5. CMS grounding

periphery of the shield to the chassis or frame where the shield has to becontinued.’Pig tail’ connection of the shield to the chassis reduces the attenuation of
the shield a high frequencies.

2.5 CMS grounding
In this section only some of the characteristics and rules for the CMS detector
grounding are presented. The goal of the chapter is not to provide a receipt to define
the detector grounding, but give some direct rules to orient the design. These rules
are not fixed and depend on the personal knowledge of the designer and grounding
philosophy chosen by the detector. This is one of the reasons why the grounding
policy must be done by only one or two persons. Assigning this responsibility to
one or more persons per sub-detector can lead to different grounding topologies,
which may direct to catastrophic consequences during the integration of thedetector. Three major rules have be considered as basic for designing the system
grounding and these are:• Grounding is not required for correct operation of the system.
• No current returns should use reference/safety ground.
• Shielding should divert pick-up currents from FEE.
The detector integration based on the electromagnetic compatibility (EMC) of thedifferent sub-systems has several stages and the grounding design is only one ofthem. The grounding design of the system constitutes only the first EMC issue inorder to achieve electrical safety and good performance for the complete detector.
The other issues, as sub-system emission and susceptibility, are oriented to control
the interference among electronic units. It is an enormous error to think that a good
grounding solves all the problems related to electromagnetic interferences.
Starting with the design of the grounding system to comply with the electricalsafety, the CMS detector must follow the CERN electrical safety rules based on
European standards. They can be achieved following these steps:• Connection to ground of all exposed metallic structures and parts.
• Avoid current returns through the ground structure of the complete detector
during normal operation.
• During faults, the ground path should be able to carry safely the fault current.
This current path not only has to have the ampacity necessary to carry the fault
current but also this path has to present an impedance low enough to activate all theover-current protections properly included in the system to prevent fire during
faults.

69

2. Grounding & Shielding

The main grounding structure of the detector is the own metallic structure of thedetector together with the concrete reinforcement structure at ground level. This is
the ground reference system of the detector (SRS). All safety and protective grounds
have to be connected to these structures. Both are shown in figure 1.3 in green color.
At CMS detector we can distinguish two different areas, where the grounding
design can follow different strategies. Those areas are the ’DC-area’ around thedetector itself and the ’400Hz distribution area’. The first area includes very low
voltage units, in the range of 2.5V to 10V with current capabilities up to 100A, andhigh voltage units ranging between 600V and 10KV with very tinny current
capability. The ’400Hz distribution area’ is composed by the three phase 400V
power distribution system with a possible installed apparent power of 500-750VA.
As it was described briefly in chapter 1, the ’DC-area’ is composed by all thefront-end electronics (FEE) of the sub-detectors. They can be arranged in millions ofchannels where the sensitive detector is close integrated to the front-end amplifierand signal processor or can be composed by amplifiers that process tinny signals
coming from chambers distributed in a vast area of the detector (muon system). In
both cases, the metallic structures, boxes or frames containing the electronics andthe detection devices have to be connected to the metallic structure of the detector
or SRS. The general practice is to use the metallic frame or box as shield or screen
for the sensitive electronics, forcing to connect internally the common point of theelectronics to the box or frame. Then, when the FEE is considered as a load, each
unit has one of the power terminals connected to the metallic box or frame, which isdifferent to the scheme represented in figures 2.2 and 2.3. Additionally, due to thevoltage levels and current capabilities associated to the equipment in this area, an
independent conductor for PE is not included and each FEE is directly grounded to
the SRS. This topology is similar to the multi-point grounding connection depicted
in figure 2.7, but the PE o green cable is not included.
Power supplies (PS) for the FEE system, in general, are located outside of thedetector, in the CMS case, they will be placed in two areas, one around 20-40 mts
away from the FEE and the other about 100-120 m. To avoid the return current fromthe FEE uses as path the SRS structure of the detector, the outputs of the power
supplies are generally floating respect to their frame or case. This frame or case has
to be grounded locally to the SRS. This topology is possible in the case of low
voltage distributions due to the extreme low voltage involved (V<45V). For highvoltage power supplies, the output terminals are not completely floating as the low
voltage PS and usually resistors or back to back diodes are connected between thecommon terminal and the PS case. This topology assumes that all the power
supplies include a transformer with screen that separates electrically the input
terminals from the output terminal, which is in general observed for all practicalapplications.

70

2.5. CMS grounding

Considering the system at the input terminals of the power supplies as individualloads, each one is now in close agreement with the diagram depicted in figures. 2.2
and 2.3. The AC or DC distribution system at this side is perfectly defined bystandards [25] and no violations of safety rules are allowed. In the ’400Hz
distribution area’, the PEs run in parallel with the power cables and they areconnected together at the secondary star of power distribution transformer. This isshown in figure 2.11, where the proposed grounding topology of the HCAL
sub-detector along the power supply is shown.

ConduitConduit

GND
Generator

PE

PE

Distribution
Transformer

GND - detector

Distribution
Panel
FEE

Conduit

ShieldPE

Distribution
Panel

AC\DC converter

GND - detector
GND - detector

Figure 2.11: HCAL grounding scheme.
The isolation transformer included in the distribution isolates in common mode, thedifferent parts of the system at low frequencies, while this isolation is degraded athigh frequency due to the parasitic components of the transformer. The use of thistransformer is still not clear, as the system will be isolated from the AC mains (thesystem will use a set of motor-generator to power the detector or static converters
with transformers). The final selection will depend on the definition of the final
configuration for the 400 Hz distribution system. The transformer in the AC-DC
converter is included to reduce the voltage (400V) distribution to extra low voltage(48V or 10V). These transformers must be able to work into a magnetic field andalso isolate the AC-LV area from the DC-extra low voltage area. These transformers
should include electrostatic screen, which will improve the rejection of low
frequency common mode currents coming from the AC line. The transformer
metallic structures and the screen have to be connected to ground.
That overview of the grounding system for CMS basically covers the first two
points described above about electric safety. The current capability of all the fault
current paths for the ’400Hz distribution area’ is well described in standards[14][30]. When the design is extended to the ’DC part’, special care must be
observed in the analysis of the fault path, particularly, its current capability and
impedance. When one of the input terminals on the FEE is energized during a fault,
the current will flow back to the source using the SRS path. This path is established71

2. Grounding & Shielding

through the connection between the FEE and the metallic case holding it. This pointis critical due to the front-end electronics in general uses the frame or case as
screening. From safety point of view, this connection is better at the input power
terminal to avoid that the fault current uses as path the FEE where the fault current
capability is probably not enough. From performance point of view, it is convenient
to connect the FEE to the screen as close as possible to the sensitive amplifier or
detector to reduce the parasitic currents coupled to the amplifier. A trade-off
between performance and safety occurs in the selection of the connection point
between the FEE and the metallic case, prevailing the design based on electricalsafety.
The brief description of grounding design based on electrical safety considers a low
frequency range, covering from DC to a few KHz. Grounding can affect theperformance of the equipment installed and it is necessary to evaluate its impact
considering the high frequency behavior of the system. As it was described inchapter 1, data signals from the FEE installed into the detector are transmitted to thecounting room via optical links. Slow control signals are electrically isolated via
opto couplers. The signal transmission via optical links simplifies considerably thegrounding configuration and allows focusing the grounding issues of the detector
to the power distribution system. This configuration allows considering the only
mechanism of interference among FEE units installed into the detector isconduction and radiation through the power cables.
The electronic load equipment installed on the detector is multipoint grounded tothe signal reference structure. Basically, reference terminal or common points of thedigital and analog electronics are connected at a convenient point at the board levelto optimize performance. Electronics, hardware and power parts are grounded
separately at module level and then connected together at a single point, which
represents the main ground per sub-detector. As it was pointed before theseconnection is a trade-off between safety and performance.
The noise generated by the PS units can be galvanically coupled to the FEE. This
noise is more severe in switched mode power supplies due the high frequency
switching of semiconductor devices. This noise has to be filtered by EMI filters
included at the input and output of the power supplies. Additionally, EMI filters
can be included at the input terminals of the FEE. Although, CM currents can flow
through the power cables, the return path for these currents will be the SRS if noshield is used in the cable, increasing the capability of the cables to radiate. Shields
reduce the radiation area for these currents and reduce the noise current flowing
through the ground of the detector.
The shield of power cables are connected to one end at low frequency (FEE side)and to both ends at high frequency. This can be achieved grounding the shield to thePS side through a capacitor. Cable’s shields connected at both side at high frequency
adds screening to the conductor from external electromagnetic fields, avoiding the72

2.5. CMS grounding

induced current to flow trough the sensitive parts of the front-end electronics. All
auxiliary equipment as cooling system, etc. should be bonded to the signal reference
system. The equipment should be isolated from the FEE to avoid antenna behavior.
Each piece of electronic equipment has to be connected to the SRS. The bonding
connections should be as short as possible with no sharp folds or bends. Flexible
and multiple straps are preferred instead of round conductors. The main goal ofthese connections is to decrease as much as possible the ground connection
impedance as well as to increase the resonant frequency of the strap. As a result, theperformance of the FEE is not affected. As example of strap effects, the groundconnection between the read-out box (RBX) to a reference copper plane has beenstudied. The HCAL-RBX is connected during the FEE immunity test via a strap as it
is shown in figure 2.12 and the inductance measured. Results are shown in table 2.1

Strap

RBX

Copper plane

Figure 2.12: RBX - SRS (copper plane) single strap connection

mA – StrapmV – StrapL

At 5 MHz

3.384

22.4

0.209 µ H

At 5 MHz

1.298

8.40

0.205 µ H

At 10 MHz

0.987

12.2

0.165 µ H

At 10 MHz

1.196

13.90

0.187 µ H

Average

0.191 µ H

Table 2.1 Strap’s inductance of RBX prototype.
The ground connection presents two resonant frequencies at 18.5 MHz and at 21.7
MHz, associated with the strap inductance and the parasitic capacitance between
the RBX to copper plane. Based on equation of figure 2.9 values for stray73

2. Grounding & Shielding

capacitances associated to the strap and to the RBX are estimated. These values
correspond to 380 pF and 275 pF. The problem associated to the strap resonancefrequency may be easily solved by decreasing the length of the straps and adding
more connections to send the resonance frequency beyond the frequency of interestof the FEE. Also these straps or ground connections should be connected to
opposing corners of the equipment and to the nearest, but separate points, on thesignal reference grids. This is shown in figure 2.13.

Strap 1

Strap 2
Strap 3Figure 2.13: RBX - SRS (copper plane) multiple strap connection.

2.6 Summary
Grounding is necessary for safety, protection and performance reasons. It is basedon standards and no violation of safety rules can be done to improve theperformance of the electronics. The metal parts of equipment, enclosures and racks,
which are susceptible to be energized by electrical currents must be grounded for
personnel safety reasons and equipment protection. No operational currents mustflow in this ground (structures and cable). The ground conductor or protective earth
must run in parallel with the phase and neutral conductors.
There are mainly two different ways of grounding the electronic equipment, they
are: the isolated ground and the standard ground. The main difference between
both grounding configurations is that in the isolated ground configuration, themetallic parts holding the load are grounded independently from the rest of themetallic parts of the system, whereas in the standard ground, the metallic box
holding the load is directly connected to the general structure of the ground system.
The standard ground configuration only needs a simple cable as ground conductor,
which is called protected earth. However, the other configuration needs one

74

2.6. Summary

additional cable, which must fulfill with the typical requirements of the ground
conductors, and is used to ground independently some areas of the system.
In cases when two or more components of an interconnected system are installed in
an area where there is a physical space between them and across their separation
data input/output cables and inter-unit power circuit cables (DC, AC or both) are
routed, there are indirect bounding problems ranging from DC to several tens of
MHz or higher that can compromise the system performance. To make compatible
the grounding with the system performance, electronic components are locallygrounded to ground planes or grids known as signal reference surfaces. The main
advantage of these structures is to provide a low impedance path for the RF noise
current and to contain electromagnetic fields between their source and the plane.
The connection layout to the equipment to the reference structures can be made indifferent ways but the two most important ones are: the single point groundconnection and multi-point ground connection. The former use only one point
where the components of the system can be grounded. For large systems, thisconnection is not recommended due to the large amount of cables needed, the large
magnitude of the impedances associated with the ground connection and the poor
CM rejection due to the parasitic components associated with the system. In themulti-point ground connection each component is grounded at two points; one atload level and other at the safety ground point. This configuration allows a muchbetter control of the noise of the system. Attention should be paid to the generationof low frequency ground loops, which could deteriorate the system performance.
The connection of the components of the system to the ground is made by straps.
These straps must be as short as possible to minimize inductive reactance of thepath. The use of at least two bond straps widely separated on the same equipment
is suggested to reduce the reactance of grounding path. These straps should havedifferent lengths.
Shielding of different components of the detectors is necessary to attenuate theelectromagnetic, electric or magnetic field that can couple into the system causing
performance deterioration or malfunction. The nature of the incident field as well as
the frequency of interest is of primary importance to select the shielding method. As
an example, magnetic materials are use to attenuate magnetic fields, whereas
conductor materials are recommended to attenuate electric and electromagnetic
fields.
The selection of shielded cables as well as the connection of the shield to theequipment is very important. The cables’s shields are made of conducting
materials, so they only can be used to attenuate electric or electromagnetic fields.
The attenuation of magnetic fields only can be performed if a current flows throughthe shield, implying the connection of the shield to ground at both ends.
CMS detector can be separated in two different parts, the AC-400 Hz distribution
area and the DC distribution area. For grounding purposes, the former is a standard75

2. Grounding & Shielding

ground system used in power distribution in which the ’protective earth’ wire runsin parallel with the system. The DC distribution area is characterized by equipment
powered by extra low voltage/high current and high voltage/very low current. The
ground system is a sub-class of the standard configuration where there is not a’protective earth’ wire running in parallel with the power distribution cables. The
detector structure is used as signal reference surface and the front-end equipment islocally grounded to this structure using straps. In this area it is necessary to takespecial care about the grounding because it is not standard topology. Groundconnections have to be established to provide, in this order, personnel safety,
equipment safety (prevent fire hazard) and optimal system performance

2.7 References[1] F. Szonsco, "Earthing of High Energy Physics Detector systems", CERN internal
document - Pre-print, 2002.
[2] F.Szonsco, "Assessment of EMC Parameters of LHC Front End Electronics", Proc. 5thWorkshop on Electronics for LHC experiments, LEB 1999, Snowmass, Colorado,
USA, pp 20-24, September 1999.
[3] P. Van der Laan, M. Van Houten, A. Van Deursen, "A grounding philosophy", Proc.
IEEE Symposium on Electromagnetic compatibility, CD-ROM Symposia Records
1955-1995.
[4] "European Standards", 93/68/CEE - (73/23/CEE), Ed. 1993[5]" CERN code C1-Electrical Code”, Ed. 1990
[6] William T. Rhoades, "Congruence of low voltage power main transient designs", Proc.
IEEE National Symposium on Electromagnetic compatibility, Denver, USA, pp285-293, May 1989
[7] Edward C. Cantwell, "Effective grounding- The key to personnel and proper
equipment operation", Proc. IEEE Symposium on Electromagnetic compatibility, pp
194-199, 1980.
[8] A. Chouvelon, W Weingarten, "Grounding as seen by TIS", TIS-GS/TM/98-01,
CERN-1998.
[9] Stanley A. Erickson, "Spacecraft electromagnetic environment prediction", Proc. IEEE
Symposium on Electromagnetic compatibility, pp 106-115, 1978.
[10] Jasper J. Goedbloed, "Electromagnetic compatibility", ISBN-0-13-249293-8, 1990
[11] Tim Williams, "EMC for product designers, Paraninfo", ISBN-0-12-55710-0, 1997.

76

2.7. References[12] Clayton R. Paul, "Introduction to Electromagnetic Compatibility", NY, Wiley
Interscience, ISBN-0-471-54927-4, 1992.
[13] "IEC 61000-5-2 Electromagnetic compatibility -Part 5: Installation and mitigation
guidelines - Section 2: Earthing and cabling", Ed. 1997.
[14] "IEEE-Std 1100- Recommended designs/installations practices - Chapter 8", Ed. 1999[15] Jhon D. M. Osburn, Donald R. J. White, "Grounding a recommendation for the
future", Proc. IEEE Symposium on Electromagnetic compatibility, pp 158-159, 1987.
[16] Reinaldo Perez, "Handbook of Electromagnetic Compatibility", Academic press,
ISBN-0-12-55710-0, 1995.
[17] Warren H. Lewis, "The use and abuse of insulated / isolated grounding", IEEE
Transactions on industry applications, Vol 25, N-26, pp 1093-1101, Nov./Dec. 1989.
[18] Henry M. Hoffart, "Single point grounding and multiple reference grounding", Proc.
IEEE Symposium on Electromagnetic compatibility, CD-ROM Symposia Records
1955-1995.
[19] G.M. Kunkel, "Practical and theoretical aspects of single point grounding", Proc.
IEEE Symposium on Electromagnetic compatibility, 1987-CD ROM Symposia
Records 1955-1995.
[20] T. Hubing, T. Van Doren and J. Drewniak, "Identifying and quantifying printed
circuit inductance", Proc. IEEE Symposium on Electromagnetic compatibility, pp
205-208, 1994.
[21] X-D Cai, G.I. Costache, "Numerical extraction of partial inductance of package
reference (power/ground) plane", Proc. IEEE Symposium on Electromagnetic
compatibility, pp 12-15, 1995.
[22] Albert Ruehli, Clayton Paul and Jan Garrett, "Inductance Calculations using
Partial Inductances and Macromodels", Proc. IEEE Symposium on Electromagnetic
compatibility, pp 23-26, 1995.
[23] F.W. Grover, “Inductance Calculations: Working Formulas and Tables”, Dover
Publications Inc., 1973.
[24] Henry Ott, "Noise reductions techniques in electronics systems", New York, NY:
Wiley, 1988, ISBN - 0-471-85068-3.
[25] Edward F. Vance, "Coupling to shield cables", 1987, ISBN - 0-89874-949-2[26] R. Schulz, V.C. Plantz and D.R. Brush, "Shielding Theory and Practice", IEEE
Transactions on EMC, Vol 30, N-3, pp. 187-201, August 1988.

77

2. Grounding & Shielding[27] L. Hasselgren and J. Luomi, "Geometrical Aspects of magnetic shielding atextremely low frequencies", IEEE Transactions on Electromagnetic Compatibility, Vol.
37, N-3, pp 409-419, August 1995.
[28] C. Taylor, C. Harrison and N. Younan, "On predicting the Effectiveness of magnetic
Shields at low frequencies", Proc. IEEE International Symposium on Electromagnetic
Compatibility, pp 176 - 178, August 1993.
[29] A. Ishiyama and H. Hirooka, "Magnetic Shielding for MRI superconducting
magnets", IEEE Transactions on Magnetics, Vol. 27, N- 2, pp.1692-1695, March 1991
[30] Thomas M. Gruzs, "Design considerations for powering 415 Hz computer systems",
IEEE Transactions on industry applications, Vol 25 N-6 pp 1102-1109, Nov./Dec.
1989.

78

Chapter 3
Conducted noise

CMS has proposed a 400Hz AC distribution system to power up the front-end
electronics installed into the detector. This system generates a set of 400V
three-phase voltage at the ground level of the CMS building and distributes it fromthat level up to the periphery of the detector in the cavern or counting room. Each
sub-detector has proposed different low-voltage (LV) units based on this primary
distribution at 400Hz. Some of them will use simple three-phase transformer andrectifier units with inductive-capacitive output filters. The transformer and rectifier
are located in the periphery of the detector and they have to operate reliably under
neutron radiation and fringe magnetic fields. In this sub-system, the final LV
regulation is achieved using linear low voltage regulators located near the front-end
electronics. Other sub-detectors have proposed to use LV units based on AC/DC
converters and DC-DC converters as post-regulators. In this equipment, the 400Hz
distribution is converted to an intermediate 48VDC distribution and DC-DC
converters perform the final step-down conversion to the required low voltages.
DC-DC converters will be placed around the detector where the environment
conditions are similar to the one described above. A simplified scheme of the
distribution system for the CMS detector is shown in figure 3.1.
All switching converters generate and emit high frequency noise. The noise
generated by DC-DC converters depends strongly on the topology of the converter,
layout design, parasitic components, etc. Emissions are mainly coupled to thesensitive front end electronics (FEE) by conduction or radiation via the cables
connected to them. To reduce the conducted noise, additional filtering is necessary
at the input and the output of the converters. These filters have to attenuate noise
components that range from the switching frequency of the converter up to 30-50
MHz.
Filters can interact adversely with the converter at low frequencies, resulting insevere performance degradation or even instability. This problem is originated by
the negative input impedance that DC-DC converters present when they operatewith tight close-loop regulation of the output voltage. This negative impedanceinteracts with the input filter, input distribution cables, and other converters79

3. Conducted noise

connected to the same distribution bus, giving place to instabilities or deteriorationof the dynamic performance. Input EMI filters have to be properly selected to avoid
this problem and also to provide the adequate attenuation in a wide range of
frequency.
The 400Hz distribution system is fully loaded by AC/DC converters. These
non-linear loads generate harmonic currents in the distribution system that interact
with the distribution system impedance and produce distortion in the sine wave
voltage. Harmonics are prejudicial as they over-heat equipment, affect the voltage
quality of the distribution, radiate noise, excite power distribution resonance,
induce malfunction of sensitive equipment, etc.
All the noise sources described and their location in the distribution system areshown in figure 3.1. This chapter addresses the analysis of the conducted noise
generated by DC-DC converters and the filter design to mitigate the noise emission.
In addition, the generation of current harmonics and their effects in the 400Hz
distribution system is briefly studied.

DC − DC
converters

FEE

AC − DC
converters

Harmonics

Stability

RF conductedemissions

400 Hz − 3φ
networkAC − DC
converters

DC − DC
converters

Figure 3.1: Power distribution layout

80

FEE

3.1. RF conducted noise

3.1 RF conducted noise
DC power distribution has been used by aerospace and telecommunicationindustries [1][2]. This topology distributes a high voltage (HV) and converts it to a
low voltage (LV) either locally or near the electronic equipment. CMS and Atlas
detectors [3][4] have proposed similar topologies to power-up the front-end
electronics. In such proposals, the primary 50Hz or 400Hz AC mains is rectified
away from the detector and DC high voltage (200V or 48 V) is distributed to theperiphery of the detector. At that location, DC-DC converters transform with highefficiency the HV into the LV required by the FEE. Those converters are locatedbetween 20-40 meters from the front-end electronics due to the intense magnetic
field that exists inside the detector.
For the CMS experiment, converters must operate reliably under high-energy
neutron radiation and fringe magnetic fields. DC-DC converters must also presenthigh efficiency, isolate galvanically the input and output, and couple a low amountof noise to the surrounding electronic equipment. Switching power converters
intrinsically generate a noise level that, in general, is not compatible with thesensitive electronics used in high energy physics (HEP) experiments. Input andoutput filters are necessary to attenuate the level of noise coupled by radiation andconduction through the cables.
Regulation of electromagnetic interference (EMI) began in early days of electronics.
Today, there exists a vast collection of standards [5][6] covering equipment used inindustry, military applications, commerce and residence. In Europe, the limits ofhigh frequency noise are specified either by generic standards (EN50081-1 forresidential, commercial, and light industry, EN50081-2 for industrial environment)
or by standards for specific product families (EN55014 for household appliances,
EN55022 for information technology equipment, or EN55011 for radio-frequency
equipment) for industrial, medical and scientific applications. In USA, the Federal
Communication Commission (FCC) issues EMC standards, with different limits forclass A and class B devices. Both FCC standards are defined for digital equipment
marketed for use in commerce, industry or business environments (class A) and aresidential environment (class B). Typically, European standards for conducted highfrequency emissions are specified in the frequency range from 150KHz to 30MHz,
and in the United States from 450KHz to 30MHz. The allowed conduction emission
levels are between 46 dBuV and 79 dBuV. These limits are imposed to the input of
the equipment under test and the compliance is verified inserting a line impedancestabilization network (LISN) between the AC mains and the power cord. The
measured values correspond to the voltage level registered across any input wire
terminated at the source with a 50 ohms impedance to ground (LISN termination).
Old Military standards for conducted emissions (MIL-STD-461 CE-03) differ fromthe other standards. It does not use the LISN1, it directly measures the emission81

3. Conducted noise

current using a current probe. Also it specifies that conducted emissions have to bemeasured on other cables in addition to the power cord. The range of frequencies
covered is between 14 kHz and 50 MHz and the emission level are between 86
dBµA and 20 dBµA . To compare these standard we should normalized themeasurement to dBµA or dBµV assuming a normalized impedance of 50 ohms.
Figure 3.2 depicts the three standards normalized in dBµV .

Figure 3.2: Conducted emission level fixed by EU, FCC and MIL standards

3.2 Switching mode power supplies
Large amount of conducted noise, in the detector, is created by the switching mode
power supplies (SMPS) installed either in the control room or on the periphery ofthe detector. SMPSs generate high frequency noise due to the switching action.
There is an important effort to reduce the volume of the power converters keeping
constant their efficiency. This improvement is achieved by increasing the switchingfrequency of the converter and using soft-switching commutation. Although a big
effort is put in the converter design to reduce the noise emission, the levels achieved
still are not compatible directly with the levels required by HEP detectors.
The conducted noise studies are carried out using VICOR power converters. These
switching power units are Zero Current Switching (ZCS) Forward Quasi-Resonant
converters (QRC) with secondary-side resonance operating in half-wave mode.
Figure 3.3 shows a simplified electrical diagram of the VICOR DC-DC converters
used during this test. This topology is a forward converter switching at zerocurrent. During each cycle, a fix amount of energy is transferred between the input
to the output of the converter. The output voltage is regulated by a closed loop,
adjusting the switching frequency of the converter, according to external conditions
1 New Military standards use a LISN
82

3.2. Switching mode power supplies

as input voltage, output current. The high efficiency of DC-DC converters operatingwith ZCS allows operation with relatively high frequencies (up to 1 MHz).

Figure 3.3: VICOR DC-DC converter scheme.
As conclusion, the main characteristics of ZCS QRC converters are:
• Reduced switching losses.
• Increased power density, since the size and weight of the magnetic components
is decreased by increasing the operating frequency.
• Reduced Electromagnetic Interference (EMI) due to ZCS.
• High reliability.
• Fast transient response

3.2.1 Conducted RF emission of SMPS
The input and output currents in SMPS contain not only the DC components that
contribute to the real power transfer, but also contain a large amount of harmoniccomponents of the switching frequency. These harmonic components are theelectromagnetic interference conducted out from the power supply through the
input and output cables [8][9].
Both the input and the output terminals of the switching power supply can beconsidered as a two terminal (positive, negative terminals) port with a common
return (chassis or case), where the terminal current can be decomposed into twomodes:• Differential mode (DM)• Common mode (CM)
Figure 3.4 shows both common and differential modes of propagation. These modes
are related with the terminal currents as:83

3. Conducted noiseIˆ+ – Iˆ- = Iˆ DM(3.1)

Iˆ+ + Iˆ- = ˆI CM(3.2)

where I+ and I- are the current at positive and negative terminals, respectively.

Icm

Id m

Vdm
V cm

Icm

Id m

Figure 3.4 CM and DM noise propagation
The propagation of the differential mode EMI takes place between conductor pairs,
which form a conventional return circuit (e.g. negative/positive conductors, line
phase/neutral conductors). The DM EMI is the direct result of the fundamental
operation of the switching converter. The DM is caused by the voltage induced
across the equivalent series inductance and resistance of input and output filter
capacitors at high di/dt. Selecting the adequate capacitors with low series
inductance and series resistance, it is possible to decrease the DM emissions.
The propagation of the common mode EMI takes place between a group of
conductors and either ground or another group of conductors. The path for the CM
EMI often includes parasitic capacitive or inductive coupling. The origin of the CM
EMI is either magnetic or electric. EMI is electrically generated when a circuit withlarge dv/dt has a significant parasitic capacitance to ground. Magneticallygenerated CM EMI appears when a circuit loop with large di/dt in it has significant
mutual coupling to a group of nearby conductors.
The CM noise is difficult to cancel. Signals with high dv/dt coupled through theparasitic capacitances between the switching devices and the heat sink and between
primary and secondary windings of the transformer induce common-mode
currents. These currents flow through the wires at the input and output of theconverter interfering with the neighborhood due to conducted and radiated noise.
In switching power converters, the same fundamental mechanism that is
responsible for conducted EMI can also generate radiated EMI. Metal cases aroundthe converter tend to attenuate the internal high frequency electromagnetic fields.

843.2. Switching mode power supplies

Input and output cables or improperly grounded apparatus can still lead to
substantial radiation.

3.2.2 Conducted noise measurements
In the HEP community there has not been a systematic approach in defining theemission levels of EMI signals [7]. Some experiments have chosen to purchaseequipment that comply with either European or American standards, but there
does not exist any quantitative limit in the emission level of power distribution andsignal cables routed inside the detectors. CMS is trying to define limits for emissionof the electronic equipment to be installed in the experiment. These limits are
established basing on a combination of emission and immunity tests, thus it isnecessary to define a complementary test for both applications. For that purpose,the values selected to measure the emission of the PS units are noise currents, asthese parameters will be used to identify the immunity of the FEE. This normalized
test measures the input and output emissions with a current probe and a spectrum
analyzer. The current measured at the positive terminal is referred to 50 ohms tocompare with any of the standardize levels.
The test layout is very important to ensure the validity of this test. The test layout
should represent the worst case in terms of current emissions, but following as
similar as possible the final topology of the PS system. For that purpose the DC-DC
converters are connected to the load and primary source in such way that creates avery low impedance path for the CM and DM currents, resulting the measured
noise currents with this test equal or bigger than the final ones presented in theexperiment. The selection of this topology introduces a safety margin in themeasurements. Also, this layout is very useful for the evaluation of the EMI filters
because the performance of a filter with very low impedance loads is smaller than
50 or 150 ohms loads. Therefore, these tests introduce a new safety margin when the
PS units are tested with filters.
3.2.2.1 Experimental set-up
To measure the input and output noise of the VICOR DC-DC converters, a set-up
was implemented consisting of a primary power supply, the unit under test and aload. An electric diagram of such a set-up is depicted in figure 3.5. A power supply
unit DELTA - SM3000 is used as primary source and the load is a resistive one. The
DC-DC power converter V300B12C250A3 operated at 200V input voltage, 7.5V
output voltage and 7.5A output current. The currents are measured by a current
probe (EMCO-Model 91550)1 and a spectrum analyzer (HP 8594E) in peak mode.

1 See appendix C - Transfer impedance curve.

85

3. Conducted noise

Two currents are recorded at the input and output terminals, the common modecurrent and the current flowing through the positive terminal. Shielded cables
connect the power converter unit to the primary power supply and the load toinclude a return path to the CM currents [8]. Common mode filter capacitors areadded between the converter’s terminals and the case and also at the input of the
primary power supply.

Heatsink

PS

Iin

Cv = 10nf

Cv = 10nf

Cload = 1µIcm-inCs = 0.22µCs = 0.22µIcm-out
Iout

Cs = 0.22µ200 V
Cv = 10nf

Cv = 10nfLoad

7.5 V – 7.5 A

DC-DCConverter
Figure 3.5: Conducted emission test layout

All the measures are performed with the spectrum analyzer in two steps, one
covering a frequency range between 10kHz and 5MHz using a bandwidth of 1 kHz
to obtain a good resolution and the other covering the range between 5 MHz and 30
MHz with a bandwidth of 30kHz. This bandwidth is adequate to obtain a goodresolution reducing the time of measurement. As an example, figure 3.6 shows thecurrent spectrum measured directly at the input of the DC-DC converter.

Figure 3.6: Noise in the positive lead86

3.2. Switching mode power supplies

For data analysis, the acquired values are converted to dBµV after correcting themeasured current by the transfer impedance curve of the current probe (Zt). All thecurrents are converted to voltage using 50 Ω as normalized impedance to compare
its magnitude with the voltage magnitude defined by standards. The data collected
is measured with the spectrum analyzer is peak mode, while the standards define theemission levels in quasi peak mode, given similar results for stationary loads. The
transformation from current to voltage follows the equations. 3.3-3.5.
I ( dBµA ) dmcm = V ( dBµV ) dmcm – Zt

I ( µA ) dmcm = 10

I ( dBµA )dmcm⎞⎛ -------------------------------⎝⎠
20( I ( µA ) dmcm ) ⋅ 50
V ref ( 50Ω ) ( dBµV ) dmcm = 20 ⋅ log ⎛ -----------------------------------------⎞⎝⎠
1 ( µV )

(3.3)

(3.4)

(3.5)

Figure 3.7 shows the noise floor at the input and the output of the DC-DC converter
when the unit is switched off. Under this conditions the DELTA PS unit is at 150 V
and 0 A.

Figure 3.7: Input and output noise floor87

3. Conducted noise

3.2.2.2 Measurement results
Figures. 3.8 and 3.9 show the measured current noise at the positive input terminal
and the input common mode current noise, respectively, for the Vicor converter
V300B12C250AL operating at Vin=200V, Vout = 7.5V and Iout = 7.5A.

Figure 3.8: Input noise, positive terminal current.

Figure 3.9: Input noise, CM current.
The filter has to attenuate the CM and DM mode components of the current tocomply with the noise specifications. The filter has to be characterized and designed
considering that both CM and DM currents are present at the input of the filter, thenit is important to analyze the CM and DM contents of the input current [5][6]. The
common mode component is directly measured and its magnitude it comparablewith the total current measured at the positive terminal in the frequency range
between 10 kHz and 50 MHz. In this frequency range, the input current is mainly883.2. Switching mode power supplies

composed by DM and CM currents and a filter with good attenuation for bothmodes is necessary. At low frequency (up to 1 MHz.), the differential mode
components are dominant and the external filter has to present strong attenuation
for the DM mode. It is interesting to point-out that if a simple DM filter is used toattenuate the noise spectrum depicted in figure 3.8, the result, after filtering, will besimilar to the noise spectrum depicted in figure 3.9. The common mode components
remain unaffected and the system does not comply with the standard.
Figures 3.10 and 3.11 show the current noise of the positive lead and the common
mode current at the output of the VICOR converter.

Figure 3.10: Output noise, positive terminal current.

Figure 3.11: Output noise, CM current.
At the output of the DC-DC converter, the noise spectrum is slightly different to theone at the input. The CM and DM emissions are very similar, and the contributionof the CM is higher than the emission at the input. This is something expected, as89

3. Conducted noise

the manufacturers pay more attention to the input as it is this area that is specified
by the standards.

3.3 Filter selection
Additional filtering is necessary at the input and output of the converter to reducethe conducted noise to values lower than the levels specified by the standards.
Filters have to provide attenuation in a wide range of frequencies covering from theswitching frequency up to 30-50 MHz. To fulfill these requirements, cascades of
selective low-pass filters are used.
The power converter includes two internal filters, one at the input and the other atthe output as depicted in figure 3.3. These filters contribute to the DM noise
attenuation but based on previous measurements, the conducted noise in both portsof the converter does not comply with the standards. To further attenuate both the
input and output conducted noise, high frequency (HF) filters are added to both
ports of the converter. This HF filter is in cascade with the internal filter included
with the converter and it is mainly designed to attenuate both DM and CM noise
components in high frequency. The complete filter at the input can interact
adversely with the converter at low frequency, resulting in severe performancedegradation or even instability of the system. A brief review about this problem ispresented in section 3.4.

Figure 3.12: Single phase filter - EMI structure
The typical structure of the HF filter added to the input and output ports of theconverter is depicted in figure 3.12. This filter attenuates both CM and DM noise
components. The CM components are attenuated by the common mode choke (L)and Cy capacitors, while the DM components are attenuated by the Cx capacitors
and the parasitic leakage inductance of the CM choke. The filter shown in figure
3.12 is a commercial unit used also in AC applications. In this case, the Cy
capacitors are connected to the safety conductor and its capacitance is limited bysafety requirements in order to avoid AC high leakage currents through the safety903.3. Filter selection

conductor. In our application, external capacitors are added in parallel with Cy andCx to achieve the attenuation required.
Manufacturers specify the insertion loss of these HF filters for DM and CM
components covering a frequency range up to 30MHz. This information allows
understanding the effect in the attenuation of the parasitic elements in eachcomponent [5].

3.3.1 Non ideal behavior of components.
Typical circuit components used in the design of passive EMI filters are discussed inthis section. Basically, the mathematical model of the components, including thenon-ideal behavior, is presented. The parasitic capacitance, resistance andinductance as well as the variation of the permeability and magnetic losses of themagnetic material are taken into account due to their influence in the filter
attenuation.
3.3.1.1 Resistor
The ratio between the voltage and current in frequency domain of a resistor has amagnitude equal to the resistance value and a phase angle equal to 0. However athigh frequency, the parasitic inductance and capacitance change considerably theideal behavior of the resistors. The model of a resistor valid in a wide frequency
range is depicted in figure 3.13.a, while the frequency response of the magnitude ofthis component is shown in figure 3.13.b. for a carbon resistance of 1.05 kΩ At low
frequency, the magnitude of the impedance is coincident with the resistancemagnitude but at high frequency the parasitic components are dominant in thefrequency behavior.

ab

Figure 3.13: HF resistance model91

3. Conducted noise

3.3.1.2 Capacitors
There are numerous type of capacitors according to the dielectric used. Dielectrics
generally used are: ceramics, polyester, tantalum, Aluminum oxide, etc. The ratio
between the voltage and current in frequency domain of a ideal capacitor presents amagnitude which decreases with the frequency at 20 dB per decade and the phase isconstant and equal to 90 degrees for all frequencies. However the parasitic
components associated with the capacitor produce important modifications in thecapacitor performance.
The capacitor can be considered as a pair parallel plates separated by a dielectricmaterial and characterized by a capacitance C. The losses in the dielectric can be
represented as a parallel resistance R diel . Generally this value is relative high. The
resistance of the plate is represented by a series resistor R s . Leads attached to thecapacitor have an inductance and a capacitance represented by C lead and L lead .
These components depend on the configuration of the leads. In general, R diel islarge enough that can be neglected and C lead is negligible respect to the intrinsic
capacitance C. Thus, the equivalent circuit for the capacitor valid in a wide
frequency range consists of the series combination of the intrinsic capacitor C, thelead inductor L lead , and the series resistor R s . This model is shown in figure 3.14a,while figure 3.14b shows the frequency response of this component for a ceramic
capacitor resistance of 470 pf .

ab

Figure 3.14: HF capacitance model
Based on this model, the magnitude of the capacitor’s impedance decreases
inversely proportional with the frequency until the frequency equals to theself-resonant frequency of the capacitor.

923.3. Filter selection

1
f o = ---------------------------------------2 ⋅ π ⋅ L lead ⋅ C(3.6)

This self-resonant frequency varies depending on the construction of the capacitor.
It can range from a few kHz for aluminum oxide electrolytic capacitors to several
MHz for surface mount ceramic capacitors.
3.3.1.3 Inductors
The ratio between the voltage and current in frequency domain of an ideal inductor
is defined by a magnitude that increases with the frequency at 20 dB per decade anda phase that is constant and equal to -90 degrees for all frequencies. However, theparasitic components [11][12] associated with the inductor modify this ideal
behavior mainly at high frequencies. For filter applications in DC-DC converters,
the inductor has to carry DC currents and also to be compact. Ferromagnetic coreswith small air-gap are used to fulfill both requirements.
The ferromagnetic core makes the magnitude of the inductor dependent of the DC
current. In an air-gapped devices properly designed, the core saturation reduces theintrinsic inductance only at high currents. In addition, the core introduces magnetic
looses [13] that can be modeled by a non-linear dissipative element in parallel with
the inductor, that takes into account the loss variation with the frequency. The series
resistance of the coil and the magnitude of the inductance change due the skin effectwith the frequency. The parasitic inter-winding capacitance reduces the impedance
presented by the inductor at high frequencies. A simple model for the inductor isshown in figure 3.15a, which consists of the combination of the intrinsic inductanceL, the parasitic capacitance C par , the series resistance R ser and a parallel resistor
R par to model the losses in the magnetic core. Figure 3.15b shows the frequency
response of this component, where all the components are assumed frequency
independent.

abFigure 3.15: HF inductor model93

3. Conducted noise

3.3.1.4 Common mode chokes
This component, similar to a high frequency transformer, consists of a pair of coils
wound around a ferromagnetic core in such a way that presents a high impedance
for CM currents and low impedance for DM currents. Figure 3.16 shows the circuit
and impedances for both modes of propagation of a ideal choke.

Iˆ1

Iˆ 2

Iˆ cm

L + M

M

Iˆ cm

L + M

L

Iˆ dm

L− M

Iˆ dm

L− M

L

Figure 3.16: CM choke equivalent impedances
If the current is decomposed in both common and differential mode, the impedancefor each mode is:
Z CM = jω ( L + M )(3.7)

Z DM = jω ( L – M )(3.8)

where L and M are the self and mutual inductance of the choke.
The parasitic components associated with the choke limit the attenuation to CM
currents introduced by this component in a circuit. The two main parasitic
components are:• Inter-windings and inter-layer capacitances• Core losses due to Eddy-currents and magnetic hysteresis.
Inter-winding and inter-layer capacitances reduce the effective Z cm at highfrequency because this parasitic effect manifests as a capacitor in parallel with Z cm .
The core losses appears as a dissipative element also in parallel with Z cm . The
effect in the circuit of this dissipative element depends on the core magnetic
material [14], the operation frequency and the input-output voltage across thechoke. Several papers in the literature [15]-[19] have reported different models of
the CM choke. The model chosen for the present analysis is described in [20] and it
is shown in figure 3.17. In this model, Lk ck represents the choke’s leakage
94

3.3. Filter selection

inductance or L dm , the Cin ck the inter-coil capacitances, Cw ck the inter-winding
capacitance, R ck the winding resistance and Rcck represents the core losses, which
depends on the magnetic material, the frequency and the input-output voltage.

Cw ck
Rc ck
R ck

Lk ck

L

k = 1
R ck

Lk ck

Cin ck

L
Rc ck

Cw ck

Figure 3.17: CM choke equivalent circuit.

3.3.2 Insertion loss
EMI filter manufacturers characterize the attenuation of the filter by the insertion
loss (IL). In a filter, the insertion loss [21] is defined by the ratio between the power
consumed by the load in a reference circuit before and after the filter is inserted into
the circuit. The IL is expressed as:
P1
V1
IL = 10 ⋅ log ⎛ ------⎞ = 20 ⋅ log ⎛ ------⎞⎝ P2 ⎠⎝ V2 ⎠(3.9)

where P 1 and P 2 correspond to the power dissipated by the load and V 1 and V 2
are the voltages across the load before and after the filter is inserted, respectively, as
it is shown in figure 3.18.

Figure 3.18: Insertion loss circuit95

3. Conducted noise

The IL depends on the terminal impedances, and therefore cannot be statedindependently of these impedances. Most of the manufacturers give the insertion
loss of the filter as function of the frequency referred to 50 ohms, because this is thenormalized impedance value for emission tests (LISN impedance). However in realapplications, this impedance does represent neither the load nor the source
impedance for the filter. The value of the source/load impedances for the filter canvary from a few ohms up to a few thousands of ohms. To correct this problem somemanufacturers give the insertion loss curves of the filter measured in accordance to
CISPR 17 [22]. The standard defines two test conditions, one using normalized 50
ohm impedances as source/load impedances and the other condition using ansource impedance of 0.1 ohms and a load impedance of 100 ohms and vice versa.

3.3.3 Filter characterization
Two commercial HF filters (Schaffner FN 2010-1 and FN 2010-20) have been used in
the present analysis. Filters have been modeled including the components with theassociated parasitic elements. The component parameters haven been calculated
based on the insertion loss curves provided by the manufacturer. The filter layout,
including the parasitic elements in each component, is shown in figure 3.19.

Figure 3.19: Parasitic components of the filter
The component parameters are shown in table 3.1. These parameters have beenvalidated calculating the CM and DM insertion loss of the filter using MATLAB
simulation. In this analysis, the skin effect1 and the variation of the magnetic losses

963.3. Filter selectionin the core with the frequency are considered. The skin effect model follows thedevelopment described in appendix A, while the core loss variation with thefrequency is modeled defining two different constant resistance values for the DM
and CM circuits. Figures 3.20 and 3.21 show the IL of the filters given by the
manufacturer, referred to 50/50 Ω , and the simulated results for the input and theoutput filters, respectively. There are small differences at high frequency, but
acceptable for the present analysis.

FN2010 - 1

FN2010 - 20

CX

0.104 µF

95 nF

L-CX

350 nH

13 nH

R-CX

0.085 Ω

0.022 Ω

CY

4. 7 nF

4. 7 nF

L-CY

9 nH

13.2 nH

R-CY

0.8 Ω

0.05 Ω

R

1 MΩ

1 MΩ

L-R

10 nH

10 nH

C-R

1.2 pF1.2 pF

L ck

12.5 mH

0.6 mH

Lk ck

50 µH

1.05 µH

R ck

⎛
f ⎞
0.6 ⋅ ⎜ 1 + -----------------3⎟ Ω⎝
10 ×10 ⎠⎛f ⎞0.08 ⋅ ⎜ 1 + --------------3⎟ Ω⎝
5 ×10 ⎠Rcck

CM -135 kΩ
DM -2 kΩ2 kΩ

Cin ck

6.5 pF0.1 pFCw ck

1.5 pF

4.5 pF

Table 3.1 Components parameters of FN2010 - 1 andFN2010 - 20 filters

1 Appendix A

97

3. Conducted noise

Figure 3.20: Insertion Losses - FN 2010 - 1- Measured and simulated results.
(50/50sym=IL-DM-Sym; 50/50asym=IL-CM-Asym)

Figure 3.21: Insertion Losses - FN 2010 - 20- Measured and simulated results.
(50/50sym=IL-DM-Sym; 50/50asym=IL-CM-Asym)98

3.3. Filter selection

3.3.4 Filter performance
The methodology followed in the selection of both input and output filters consists
on measuring the conducted noise generated by the power converter at both the
input and output terminals and estimating the filter attenuation required to satisfy
some standard. Assuming the system has to comply with the European standard
level EN-55022 at both the input and the output ports, the filter attenuation
required can be estimated from figures 3.8 and 3.10. For the input port, it isnecessary an insertion loss greater than 40dB at low frequencies for DM and noise
reductions greater than 30dB in the high frequency range for both DM and CM
components. For the output, it is necessary attenuations greater than 45dB at low
frequencies for DM and noise reductions greater than 30dB in the high frequency
range for both DM and CM components.
Filter manufacturers specify the insertion loss (IL) of HF filters for DM and CM
components covering a frequency range up to 30MHz. This insertion loss generally
is referred to a load and a source impedances of 50 ohms, which generally is quite
far from realistic values. In real applications, the impedance presented to the HF
filter by the converter and the line can have values between a few ohms andthousands ohms, which result in different filter attenuations. Additionally, thereactive components of the filter can resonate with the source and the load
impedances. Some manufacturers specify the DM insertion loss referred toload/source impedances of 0.1/100 Ω and 100/0.1 Ω using the standard CISPR17[23], however this curves do not provide enough information to do the optimal
selection of the filter for the DC-DC converters.
The main problem to synthesize the HF filter is the lack of knowledge of the source
and load impedances connected to the filter terminals. These impedances define
partially, the insertion loss of the HF filter. To calculate the magnitude of the filter
components that achieve the IL required, the worst filter performance is studied bychanging the magnitude of load/source impedances in a practical range. The
load/source impedances are assumed resistive given an uncertainty due to possible
resonances between the filter impedances and the load/source impedances,respectively. Based on this study, it is possible to define the filter attenuation closer
to the real conditions. Once the filter IL for the worst condition is calculated, theconducted emission at the input and output of the DC-DC converter with HF filter
are estimated.
The HF filter design is performed following the decomposition of the filter in adifferential mode sub-circuit and a common mode sub-circuit. In order to estimate
the noise current magnitude after the filter is added to the converter, the measured
currents without filter have to be decomposed in DM and CM. This decomposition
is straight-forward when a LISN is used to measure the noise current. In our casethat currents are measured directly with a current probe, the common mode current
is directly measured and the differential mode current is estimated based on the99

3. Conducted noise

terminal current and the common mode current. The magnitude of the current isrecorded by the spectrum analyzer, resulting unknown the phase between the CM
and the terminal current. To estimate the DM current, the worst case is consideredby assuming that the DM current has a phase shift of 180 degrees respect to CM
current. Then, the DM current magnitude is calculated as the sum of both thepositive terminal current and CM current magnitudes. Having both the CM and theDM current magnitude without filter, the current after filtering is estimated by
dividing the current magnitude by the insertion loss magnitude. Once the CM andDM output current are estimated, the terminal current is calculated by adding them,assuming the worst case that both current components are in phase. During thisdecomposition the background noise shown in figure 3.7 is considered as a
non-correlated noise with the current measured. Its effect in the estimation is
included at the output adding its magnitude in quadrature to the estimated current.
3.3.4.1 Output filter performance
Figure 3.22 depicts the layout of the complete filter, showing the components thatdefine source impedance, Z S , and the load impedance, Z L , presented by theconverter and the external circuit to the HF filter.

Output stage − DC − DCconverter

C2

Output − EMI Filter C4

FEE
Cable

ZS

C1

ZL

C3

Figure 3.22: Output filter connections.
The source impedance is defined by the output capacitors of the DC-DC converter
and the CM capacitors included between the output terminals and the case. In DM,
this impedance is low up to high frequency where the influence of the parasitic
inductance of the layout becomes important. It is possible to assume this impedance
can range from m Ω in low frequency up to tenths of ohms in high frequency. For
the CM topology, the internal impedance of the converter is generally high, being
the source impedance mainly defined by the CM capacitor C2 (10 nF) and thecomponents of the output circuit layout. In general, it can be assumed that this
impedance is lower than 50 ohms above 200 kHz.

100

3.3. Filter selection

The load impedance of the filter in DM is defined by the cable that connects thefilter to the load and the input capacitors of the FEE. This impedance is generally
low at low frequency increasing with the frequency due to the inductive effect ofcable.
The HF output filter is implemented using the commercial part FN2010-20 adding
the polyester capacitor C1 of 0.33 µ F to achieve the required attenuation. In thisanalysis, the parasitic components of this capacitor (Lc=8 nH , Rc=0.01 Ω ) havebeen included. This filter can achieve a DM-IL greater than 30dB in high frequency
for a set of sources and loads.

ab

Figure 3.23: Insertion loss for the HF output filter in differential mode a) IL variation fordifferent Z S , b) IL variation for different Z L
Figure. 3.23 shows the IL of the output filter in differential mode. Part a) depicts the
IL for a fix load impedance Z L = 5 Ω and for different values of the source
impedance 0.1, 10, 50 and 100 Ω . The plot shows that the filter attenuation is notvery sensitive to source impedance variations for a given load impedance. A source
impedance change by a factor 1000 produces a maximum variation of the IL of
about 22-24 dB (factor 20). Part b) shows the filter attenuation is not influenced byvariations of the load impedance when it changes in the range between 5 and100 Ω . For lower values of the load impedance the IL increases. From figure 3.23, itis possible to conclude that the worst case corresponds to the condition Z S = 0.1 Ωand Z L = 100 Ω . This insertion loss is used to estimate the noise current attenuationin DM introduced by the complete filter, which seems to be enough to achieve therequire attenuation at high frequency.
For the CM noise, a similar procedure was followed assuming that the CM source
impedance can change between 1 Ω and 50 Ω and the CM load impedance
between 5 Ω and 150 Ω . Figure 3.24 shows the CM insertion loss for that set of101

3. Conducted noise

impedances. From this figure, it is possible to conclude that the worst casefrequency where the common mode starts to be important, corresponds to the
combination Z S = 50 Ω and Z L = 5 Ω . This attenuation is used to estimate thenoise current reduction introduced in CM by the HF filter.

ab

Figure 3.24: Insertion loss for the HF output filter in common mode. a) IL variation fordifferent Z S , b) IL variation for different Z L .
Based on the worst case CM-IL, the CM noise at the output of the converter with HF
filter is estimated. Figure 3.25 shows the estimated values of CM output current
noise for the DC-DC converter with the HF filter, while figure 3.26 depicts themeasured CM output noise current for the converter including the same HF filter.

Figure 3.25: CM output current conducted emissions of Vicor converter with HF filter
Simulated values.

1023.3. Filter selection

Figure 3.26: CM output current conducted emissions of Vicor converter with HF filter
Measured values.
The estimated and the measured noise current values have a similar spectrum.
There are two clear differences between both plots. The first difference is in thesecond, third and fourth harmonics. Measurements (figure 3.26) shows theperformance of the real filter (figure 3.25) is better in that frequency range than the
IL calculated for different load/source impedances. The origin of this difference canbe found in the different characteristics of load and source impedances between thereal and the simulated circuits. Resistive load and source impedances are used in
the simulation model. However for the real circuit, the load and source
characteristics are either inductive or capacitive. They change with frequency
producing a dynamic variation of the filter performance with the frequency. The
selection of the constant resistor magnitude for the load and source represents theworst case, allowing to limit the uncertainties created by the impedance variation
with the frequency. The second difference is found in the frequency range between
2.5 and 5 MHz. For this frequency range, the measured values are higher than theestimated ones. The origin of these differences is due the measurement set up. The
circuit layout can affect the performance of the filters to CM noise due to the longcables that connects the ground of the DC-DC and ground of the filter. This can beeasily observed adding an inductance in the ground path in the simulation model,
which produces a drop in the IL-CM of the filter.

C1

FN2010-1

VICOR

FN2010-20

C2
C1
Figure 3.27: RF conducted test layout for VICOR converter.

103

3. Conducted noise

As it is shown in figure 3.27, long cables are necessary to perform current
measurements between the DC-DC converter and filter. For the final version, thelayout can be improved by placing all the units on a ground plane, which decreases
the impedance of the grounds paths and connections improving the performance ofthe filter.
To estimate the total noise at the output of the converter after the HF filter is added,
from figures 3.23 and 3.24, it is possible to conclude the worst case attenuation
corresponds to the combination Z S = 50 Ω and Z L = 5 Ω for the CM-IL and
Z S = 0.1 Ω and Z L = 100 Ω for the DM-IL. Based on the DM-IL and the CM-IL
worst cases, the noise at the output of the converter is estimated. Figure 3.28 and3.29 show the estimated and real output noise current for the VICOR converter with
HF filter.

Figure 3.28: Output current conducted emissions of Vicor converter with HF filter.
Simulated values.

Figure 3.29: Output current conducted emissions of Vicor converter with HF filter.
Measured values.

1043.3. Filter selection

It is possible to observe a close agreement between the estimated and the measured
noise current values. Again, the estimated values are higher than the measured
ones, giving a safety margin during the selection process of the filter. Also, there is
only a significative difference between both plots in the frequency range between
2.5 and 5 MHz, as analyzed in figures. 3.27 and 3.26.
Based on the simulated results at low frequency the attenuation is not enough tocomply with the EN 55022-Class A. To improve the IL at low frequency, twodifferential mode capacitors in parallel (C3) are added to the HF filter output.
ceramic SMPS-MLC1 capacitors of 82 µ F are selected to achieve enough attenuation(45 dB) at low frequency for different load/source conditions. These capacitors arecharacterized by a very low parasitic series resistance, being very important toimprove the filter performance at low frequency. In simulations, the parasitic
components of the capacitor (Lc=2.4 nH , Rc=0.0025 Ω ) have been included.
The DM-IL for the new filter are re-calculated for different input and output
impedances. Figure 3.30 shows the insertion loss of the previous HF filter with theextra capacitor (C3) in differential mode. Part a) depicts the IL for a constant load,
while the source impedance is changed. The DM-ILs are quite invariant with thesource impedance variation. A similar characteristic presents the DM-IL when theload impedance changes, as depicted in fig. 3.30 part b).

ab

Figure 3.30: Insertion loss for the HF output filter in differential mode with the addition of
the capacitor C3. a) IL variation for different Z S , b) IL variation for different Z L
From these plots, it is possible to observe that the DM-IL worst case corresponds toa low source and a low load impedance. The DM insertion loss for Z S = 0.1 Ω and
1 Capacitor from AVX company

105

3. Conducted noise

ZL = 0.1 Ω is used to estimate the DM noise current attenuation introduced by thecomplete filter. The capacitor C3 added to the output does not affect the CM
emissions, which based on the simulated results depicted in figure 3.25 are stillslightly high. To attenuate the CM emission in the frequency range between 200
kHz and 2 MHz, two polyester capacitors (C4) of 40 nF are include between thefilter terminals and ground. These capacitors are shown in figure 3.22. The CM-IL of
the HF output filter with the capacitors C3 and C4 is shown in figure 3.31 fordifferent load and source impedances. In this simulation, the parasitic componentsof the capacitor C4 (Lc=2.4 nH , Rc=0.050 Ω ) have been included.

ab

Figure 3.31: Insertion loss for the HF output filter in common mode with the addition of thecapacitor C4. a) IL variation for different Z S , b) IL variation for different Z L .
The filter CM-IL with the CM capacitors C4 is, in general, quite invariant to source
and load impedances except for the case of large source impedances and small load
impedances. Based on the CM-IL, the worst case corresponds to Z S = 50 Ω and
Z L = 5 Ω . If we compare the IL depicted in figure 3.31 with the IL shown in figure
3.24 corresponding to the HF filter without the CM capacitor C4, it is possible tonote that CM-IL curves are practically identical at low frequency. At this frequency,
the main element defining the CM attenuation is the CM choke, that in our case is
the same in both cases. If a higher attenuation is required at low frequency, it isnecessary to select another filter with larger series inductance.
Based on the DM-IL defined for Z S = 0.1 Ω and Z L = 0.1 Ω and the CM-IL set by
Z S = 50 Ω and Z L = 5 Ω , the terminal noise current is estimated. Figure 3.32
shows the estimated noise current.

1063.3. Filter selection

Figure 3.32: (CM+DM) output conducted emissions of Vicor converter with filter plus
extra capacitors C3 and C4. Simulated values
3.3.4.2 Input filter performance
Following a procedure similar to the output filter design, the HF input filter isdesigned to comply with the EN 55022-Class A standard. Figure 3.33 shows theelectrical scheme of the DC-DC converter, the HF filter and the load.

Output stageof AC − DC
Cable

Input − EMI Filter

C1
ZL

C5

Input stage − DC − DCconverter

ZS

C2

Figure 3.33: Input filter connections
The source impedance for the DM topology is defined by the input stage of the
DC-DC converter, which basically depends on the internal inductance and CM
capacitors (C2) added to the input terminals of the Vicor converter. The magnitudeof the internal inductance is 60 µ H and the CM capacitance is 10 nF, giving aresonance frequency for the DM filter of 300 kHz. Above this frequency, the source
impedance is mainly defined by the CM capacitors and the layout of the input

107

3. Conducted noise

circuit. If the resonance between the internal inductor and the CM capacitors isdamped, the source impedance for the DM topology can be assumed lower than 200
ohms for a frequency range that extends up to 50MHz. For the CM topology, theinternal impedance of the converter is generally high, being the source impedance
mainly defined by the CM capacitors C2 and the parasitic components of the input
circuit layout. In general, it can be assumed that this impedance is lower than 50
ohms above 200 kHz.
The load impedance of the filter in DM is defined by the short cable that connects
the filter to the primary power supply and the output capacitors of the power
supply. This impedance is generally low. In case of CM, these load impedance aremainly defined by the capacitor of 0.22 µ F that connects the shield to the negative
output of the primary voltage as it is shown in figure 3.5.
The HF input filter is implemented using the commercial part FN2010-1 with theadditional capacitor C1 of 0.33 µ F to achieve the required attenuation at highfrequency. In this analysis, the parasitic components of the capacitor (Lc=8 nH ,Rc=0.006 Ω ) have been included. This filter itself can achieve a DM-IL greater than
40dB in high frequency for a set of source and load impedances.

abFigure 3.34: Insertion loss for the HF input filter in differential mode. a) IL variation fordifferent Z S , b) IL variation for different Z L .
Figure 3.34 shows the insertion loss of the HF filter in differential mode. Part a)
depicts the IL for a constant load Z L = 1 Ω , while the source impedance is changed
from 10 ohms to 200 ohms. This plot shows a relative independence of the IL respect
to the magnitude of the source impedance. Part b) depicts the IL for different load
impedances, while the source impedance is kept constant at Z S = 200 Ω . This plot108

3.3. Filter selection

shows the filter attenuation is sensitive to load variations, but the IL is not generally
lower than 40dB at medium frequency. From figure 3.34, it is possible to conclude
that the worst case corresponds to the condition Zs = 200 ohms and Zl = 0.1 ohms.
The insertion loss depicted in these plots is used to estimate the noise current
attenuation in DM introduced by the complete filter. These plots show theattenuation required at high frequency is achieved for the set of loads and sources
considered but the IL is not sufficient at low frequency.
For the CM noise, a similar procedure is used, considering that the CM source
impedance changes between 1-50 ohms and the CM load impedance between 5-150
ohms. Figure 3.35 shows the CM insertion loss for that set of impedances. From thisfigure, it is possible to conclude the worst case in the frequency where the common
mode starts to be important, corresponds to the combination Z S = 50 Ω and
Z L = 5 Ω . This attenuation is used to estimate the noise current reduction
introduced in CM by the HF filter.

ab

Figure 3.35: Insertion loss for the HF input filter in common mode. a) IL variation fordifferent Z S , b) IL variation for different Z L .
Based on the worst case IL for differential mode and common mode, the noise at the
input of the converter is estimated. Figure 3.36 shows the estimated input current
noise for the Vicor converter adding the HF filter with the capacitor C1. Figure 3.37
depicts the measured input current noise for the converter including the same HF
filter.

109

3. Conducted noise

Figure 3.36: Input current conducted emissions of Vicor converter with HF filter
Simulated values.

Figure 3.37: Input current conducted emissions of Vicor converter with HF filter
Measured values.
It is possible to observe a close agreement between the estimated and the measured
noise current values. Basically, the estimated values are higher than the measured
ones, giving a safety margin during the selection process of the filter. There is only asignificative difference between both plots in the frequency range between 2.5 and 5
Mhz. For this frequency range, the measured values are higher than the estimatedones. The origin of this difference is the measurement set up as it has been detailed
in the previous section 3.3.4.1

110

3.3. Filter selection

As it was predicted above, the attenuation is not enough to comply with the EN
55022-Class A at low frequency. A differential mode capacitor C5 has to be includedin the HF filter to improve the filter performance to comply with the standards class
A. A capacitor of 5 µ F between input terminals of the HF filter provides 40dB of IL
at low frequency for different load/source conditions. The DM-IL of the new filter
are calculated again for different input and output impedances. In this simulation,
the parasitic components of the capacitor (Lc=2.4 nH, Rc=0.150 Ω ) have been
included. Figure 3.38 shows the insertion loss of the previous HF filter with theextra capacitor (C5) in differential mode. Part a) depicts the IL for a constant load,
while the source impedance is changed. The DM-IL, are quite invariant with thesource impedance variation. However, the DM-IL results are quite sensitive to load
variations. Part b) depicts the IL for different load impedances, while the source
impedance is kept constant. The worst condition IL is used to estimate the noise
current attenuation in DM introduced by the complete filter.

ab

Figure 3.38: Insertion loss for the HF input filter in differential mode with the addition of
the capacitor C5. a) IL variation for different Z S , b) IL variation for different Z L
The capacitor C5 does not produce any variation in the CM-IL of the HF filter withthe capacitor C1, being the CM-IL exactly the same to the ones shown in figure 3.35.
Again, based on the worst case IL for differential mode and common mode, thenoise at the input of the converter is estimated. Figure 3.39 shows the estimatedinput current noise for the DC-DC converter including the new HF filter.

111

3. Conducted noise

Figure 3.39: Input current conducted emissions of Vicor converter with HF filter plus C3
Simulated values.
The level selected for the present analysis correspond to the limits at the mains
terminal of the most common specific European standards EN 55011 and 55022. The
final level selection for the emission test will be based on the FEE immunity test,
which will be presented in future sections. In case of selection of a different level
defined by other standards like military, FCC or a new one defined by HEP, thedesign methodology does not change. Only the limit levels will be changed.

3.4 Stability problems with SMPS - Negative
input impedance
DC-DC switching converters with tight output voltage regulation operate as aconstant power load. The instantaneous value of the input impedance is positive
but the incremental or dynamic impedance is negative. Due to this negative input
impedance characteristic interaction among switching converters and another partof the system connected to the same distribution bus may result in systeminstability [24][25][26].
To analyze the behavior of the converter and the interaction with the rest of thesystem a reduced model of the system is necessary. The reduced model has torepresent the behavior of the system at low frequency in the range between DC andfrequencies near the bandwidth of the power converter. In this frequency range, thepower converter behaves at the input as a constant power load in cascade with the
input filter. The rest of the system can be modeled as follows: the distribution line
can be approximated by a lumped inductance in series with a resistor and the HF
filter can be reduced to the DM capacitors.

112

3.4. Stability problems with SMPS - Negative input impedance

To present a qualitative behavior of the converter at the input terminals, let us
consider first the simple equivalent circuit depicted in figure 3.40 It represents a
VICOR converter connected to a primary source with short leads. Using as statevariables, the inductor current il and the capacitor voltage V c , the state equation
is:

dV c
Pc
C ⋅ --------- = il – ----dt
Vc(3.10)dilL ⋅ ------- = E – ( il ⋅ r l ) – V cdt+
E(3.11)

L

rl

il

vc

-

C
Pc/vc

Figure 3.40: Model at the input terminals of the DC-DC converter.
This equation has two real valued equilibrium points if the condition shown inequation 3.12 is verified.
2E
r l < ------------4 ⋅ Pc

(3.12)

One of the equilibrium points is the operation point of the converter and the other isdefined for low voltage and high current, which in general, is unstable. In order toforce the converter's operation around of the equilibrium point, converters either
include limits into the dynamic range of the control circuit or disable the operationof the power transistor for low values of the input voltage. VICOR converters
disable the unit if the input voltage magnitude is outside of a voltage band aroundthe equilibrium point (e.g. Vnom=300V, Vin=180-375V). In this case, the converter
can still be modeled by equations 3.10 and 3.11 but including the condition, P c ≠ 0
if V c is between 180V and 375V, and Pc = 0 if Vc is outside of this region. Figure
3.41 shows the phase portrait of equation 3.10 and 3.11, including the last condition
for different initial conditions. Trajectories converge to the equilibrium point SS1
assuming it is stable. The stability of this point is defined by the condition:

113

3. Conducted noise

Pc ⋅ Lr l > -------------C ⋅ Vc

(3.13)

where Vc is the capacitor voltage at equilibrium.

Figure 3.41: Phase portrait of the DC-DC converter input filter
As conclusion from this brief analysis, to analyse the stability of the system, theconverter model can be simplified by a linearized model around the equilibrium
point (small-signal analysis). The region of convergence can be estimatedanalytically or by simulation using a non-linear model of the converter. The
linearized model of the converter at input terminal is characterized by a negativeE
resistance of magnitude
r n = – --- , where the E is the DC input voltage andili l is the DC input current. This current depends of the load of the power converter
and r n can take different values according to the operating conditions.
Let us consider now a generic DC power distribution system composed by one
AC/DC converter, a distribution line of 120 meters and N converter units connectedto the end-point, as it is depicted in figure 3.42. Each DC-DC converter unit iscomposed by 2 DC-DC converters, connected in parallel at the input. Only one
input HF filter is used per unit. The values of the components are shown in table 3.2

114

3.4. Stability problems with SMPS - Negative input impedance

EMI filter

120 mts . − Cable

ro

Lo
Co

rs

Ls

Rp

− R n1r1

L1 c1

DC − DC / 1

Cdr2

AC − DC converter

L2 c 2DC − DC / 2− Rn 2
Figure 3.42: Generic DC power distribution system layout.

DC-DC / 1

DC-DC / 2

Cable

L 1 = 60 µ H

L 2 = 100 µ H

L s = 75 µ H

C 1 = 350 nf

C 2 = 350 nf

r1 = 4 Ω

r2 = 4 Ω

r s = 0.4 Ω

AC-DC
unit

EMI filter

L o = 10 mH
Cd = 5 µ f

C o = 4000 µ f– R n1 = 246 Ω – R n2 = 213 Ω·
r o = 0.1 Ω

r p = 1000 Ω

Table 3.2 Parameters of generic DC power distribution
system.
At the distribution bus, the system can be represented by a simplified block
diagram, as shown in figure 3.43. The source sub-system contains the impedance of
the AC mains, the AC/DC converter and the LV distribution cable. The load
sub-system is composed by N DC-DC converter units. The source sub-system isstable when it is loaded by a resistor. Each DC-DC converter unit is stable if it isconnected directly to a power supply.

115

3. Conducted noiseE

Source sub-system

Load sub-system

Fs

Fc
Zo

Von

Zin

Figure 3.43: System block diagram.
Assuming the source sub-system has an input/output transference Fs and each
DC-DC converter a transference Fc, the overall transference is given by.

V on
Fs ⋅ Fc
Fs ⋅ Fc-------- = ---------------- = ---------------E
Zo
1 + Tm
1 + ------Z in(3.14)

where Z o is the output impedance of the source sub-system and Z in is the input
impedance of the load sub-system. Due to both Fc and F s are stable transference–1functions. The stability of the system is defined by the term ( 1 + T m )thatrepresents the loading effect between the source and load sub-systems.
If Z in » Z o for all frequencies, the loading effect is negligible. This condition canbe difficult to achieve in all the frequency range. This rule prevents any noticeableinteraction between source and load sub-systems and may be overly conservative.
If Z o is larger than Z in a considerable loading effect exists. It does not necessarily
imply a stability problem. In this case, either the Nyquist criterion or Bode based
analysis can be applied to the gain ( T m ) to determine the system stability[2][27][28][29][30].
Figure 3.44 shows, in the upper plot, the Bode plot of the output impedance of thesource sub-system and the input impedance of the load sub-system for different
capacitance Cd (figure 3.42). This capacitance is included to improve the LF noise
filtering and improve the stability in the high frequency region (around point B). In
that area, figure 3.44 shows that Tm is equal to one and the phase is near 180°. Plotsin figure 3.44 depict the load impedance for only one DC-DC converter unit
connected to the bus. With an increasing number of converters connected to the bus,
the input impedance Z in decreases, and the stability of the system becomes criticalat low frequency (point A). At this frequency, there exists interaction between the
AC/DC converter filter and the negative impedance of the DC-DC converters. In
this case, to improve the stability margin, it is necessary to increase the damping of
the AC/DC converter.

116

3.4. Stability problems with SMPS - Negative input impedance

Figure 3.44: Bode plots of Tm , Z o and Z in .
In summary the influence of the damping resistor, DM filter capacitor and numberof DC-DC units connected to the same bus bar are limited for stability reasons andthus it is necessary to achieve a compromise between these limitations. Table 3.3
summarizes for the presented example presented the limitations of the number ofunits with respect to the damping resistance and EMI filter capacitor.

Maximum number of DC-DC
units
Damping
Resistance(Rp)- Ω

EMI filter capacitor(Cd)- µ f
0.1

5

30

3

7

1000

3

5

Table 3.3 Stability summary - Maximum number of
DC-DC units1173. Conducted noise

3.5 Harmonics
A 400Hz AC system will be used as primary system to distribute power to thefront-end electronics at CMS. The topology is based on motor-generators (M-G) thatconvert the 50Hz AC mains into a three phase 400Hz, 400V system that is
distributed to the detector. There are three set of M-Gs that are connected in parallelto create the 400 Hz primary source. A back up M-G unit will be installed and will
be used as replacement of the original M-G in case of failure of any unit or during
maintenance. The proposed distribution system is depicted in figure 3.45. The
motor-generator set will be installed at the surface level of the detector building.
The power is transmitted to the counting room where it is separated in different
networks which supply power to specific areas of the periphery of the detector. At
that location, AC-DC converters transform the AC power to the DC-LV power
needed by the DC-DC converters and the LV regulators feeding the front-end
electronics.

Distr.Panel

400V/400Hz

DC − DC
converters

∆-Y

MG

FEE

AC-DC

400V/50Hz
LV/DC

∆-Y
AC-DC

∆-Y

MG

AC-DC

LV Reg

400V/50Hz
Area 2
Counting room

Area 3
Surface level

100mts.

Area 1
Periphery of thedetector

100-120 mts.

Area 0Detector

20-40mts.

Figure 3.45: 400Hz power supply distribution block diagram
The AC-DC converters are the total loads connected to the 400Hz distribution
system and they are highly non-linear. They generate harmonic currents thatinteract with the distribution system impedance and produce distortion in the sine
wave voltage. Harmonic are prejudicial because they over-heat equipment, radiatenoise, excite power distribution resonance if there exists, induce malfunction ofsensitive equipment, etc. Operating at 400Hz, the harmonics extend in a range of
frequencies higher than 50 Hz harmonics, inducing more problems [31]. In this118

3.5. Harmonics

sense, harmonic standards for 400Hz systems are more strict that standards limiting
harmonics in 50/60Hz systems.
Power generators and static converters generating a sine wave produce harmonic
voltages as a product of the conversion. Commercial generators specify a total
harmonic distortion (THD) less than 4%, individual frequency voltage harmonics
less than 3% and total deviation less than 3%, while static converters can produce an
output sine wave with a THD better than 3% for all the load range. In both cases,
the quality of the output waveform is very high.
It will be ideal to compensate harmonics at the point where they are generated, but
in-situ compensation is not economic and it is more practical to force to the loads toreduce the number of harmonics injected to the distribution system. The definitionof harmonic indices and limits are characterized by [31];• The values given by the harmonic indices should be physically meaningful andstrongly correlated to the severity of the harmonic effects.
• It should be possible to determine by measurements whether or not the limits of
the harmonics indices are met.
• Harmonic indices should be simple and practical so that they can be widely usedwith ease.
Harmonic effects differ substantially depending on the characteristics of theequipment affected. Therefore, the severity of the harmonic effects imposed on all
types of equipment cannot be perfectly correlated to a few, simple indices.
Recommended harmonic indices are: Individual and total voltage distortion,individual and total current distortion and depth of total notch area and distortionof bus voltage distorted by commutation notches.
The IEEE Std 519-1992 [32] describes the recommended practice to define harmonic
limits for individual consumers at 50/60Hz. This standard defines the limits forindividual loads based on the load size, load diversity, etc. The objectives of theharmonic current limits are to limit the maximum individual frequency voltage
harmonic to 3% of the fundamental and the voltage THD to 5%. Current limits
based only on voltage distortion are not complete because it assumes the unlimited
ability of the generator and distribution system to absorb harmonic currents [33].
This standard defines the absolute harmonic current limits using similar criteria
than above. For dedicated systems, as CMS, it limits the individual harmonic
current for low order harmonics to 4%.
To estimate the effect of the load harmonic current, different configurations of threephase rectifiers coupled with a step-down transformer are used. For the firstestimation, the transformer is assumed ideal and the effect of the switchingoverlapping of the rectifiers are neglected. Three phase rectifiers coupled bytransformers have the advantage of using the winding connection at both theprimary and secondary to cancel harmonics.

1193. Conducted noise

The topology of a three-phase full-bridge rectifier with a Delta/Wye transformer isdepicted in figure 3.46. Line-line and line neutral voltages and ideal line current
waveforms are shown in figure 3.47 for this converter. The harmonic current levels
and the voltage drop of those harmonic currents in the distribution lines are
depicted in figure 3.46. This analysis was carried out assuming a distribution linewith a size wire AWG#7with and a length of 100mts. At the end of the line, 15
power converters are connected.

I1

I2

Figure 3.46: Electric scheme of the AC/DC converter

Figure 3.47: a) Primary and secondary phase current and voltage. b) Spectrum of therectifier input current, Spectrum of the voltage drop in the distribution lines
From figure 3.47, the individual frequency voltage harmonic is about 2.3% that isclose to the limit defined by the IEEE-Std 519. The individual frequency current
limit defined by the same standard is 4% of the fundamental for harmonic orders
lower than 11, and the converter analyzed has the 5th harmonic component equal to
20% of the fundamental while the 7th harmonic component is 14.3%. If some of the120

3.6. Summary

limits presented here are going to be used in the 400Hz distribution, particular caremust be taken to reduce the harmonic content produced by the load using filteringor harmonic cancellation.

3.6 Summary
The CMS LV distribution system is composed by switching power supplies and
AC-DC converters that generates conducted electromagnetic emissions andharmonics in the distribution cables. All switching converters generate highfrequency conducted noise, which depends strongly on the topology and parasitic
components of the converter. The measurement of this noise current is very
important to evaluate the performance of the power supplies. Standards generally
use a LISN to perform these measurements. This device presents a constant
impedance of 50 ohms to the noise sources in certain frequency range. However, in
the real system, DC-DC converters are connected directly to the load or primary
power source using cables and the impedance presented by the system to the input
or output of the converter varies from a hundreds of mohms to hundreds of ohms.
This configurations may generate higher conducted emission than theconfigurations loaded with 50 ohms. For that purpose, noise emissions aremeasured based on a topology closer to the real one where the converter load is low
impedance.
The conducted emission can affect the performance of the FEE, being necessary theuse of EMI filters to attenuate these emissions to acceptable levels required to FEE.
Commercial filters are used to attenuate these emissions. The filter performance isdefined by the insertion loss of the filter. The IL depends strongly on the
impedances to which they are connected, this impedances can change with thefrequency or resonate with the filter reducing the filter performance. The
differential mode-IL is more sensitive to impedance variations than the common
mode-IL, being necessary to select the adequate load and source impedance of thefilter that allows to estimate the filter performance with enough safety margin.
Input filters interact adversely with DC-DC converters and the distribution system
at low frequency, resulting in severe performance degradation or even instability.
This problem is originated by the negative input impedance that DC-DC converters
present when they operate with tight close-loop regulation of the output voltage.
These instability limits the number of DC-DC converters connected to the same
AC-DC box. It can be controlled establishing a compromise between damping,differential mode capacitor and number of units connected to the same bus.
The 400Hz power distribution system of CMS is fully loaded by non-linear loads.
These non-linear loads generate harmonic currents in the distribution system that
disturbs the voltage generated by the motor-generators sets. The harmonics can be121

3. Conducted noise

cancelled forcing to the loads to reduce the number of harmonics injected to the
distribution system.

3.7 References[1] P.Lindman, L. Thorsell, "Applying Distributed Power Modules in Telecom Systems",
IEEE Transactions on Power Electronics, Vol 11, N-2, pp 365-373, March 1996.
[2] B. Cho, F. Lee, "Modeling and Analysis of Spacecraft Power Systems", IEEE
Transactions on Power Electronics, Vol 3, N-1, pp 44-54, January 1988.
[3] J. Kierstead, H. Takai, "Switching Power Supply Technology for ATLAS Lar
Calorimeter", Proc. 6th Workshop on Electronics for LHC experiments, pp 380-382,
September 2000.
[4] B. Allongue et al., "Design Consideration of low Voltage DC Power Distribution",
Proc. 6th Workshop on Electronics for LHC experiments, pp 388-392, September
2000.
[5] C.R. Paul, "Introduction to Electromagnetic Compatibility" NY: Wiley-Interscience,
1992, ISBN-0-471-54927-4[6] T. Williams, "EMC for Product Designers", 1997, ISBN-0-7506-2466-3[7] F.Szoncso, "EMC in High Energy Physics", Proc. 5th Workshop on Electronics for
LHC Experiments, LEB 99, pp 20-24, September 1999
[8] F. Arteche, C. Rivetta, F. Szoncso, "Effects of CM and DM noise propagation in LV
distribution cables", Proc. 9th Workshop on Electronics for LHC experiments,
LECC2003, pp 380-385, Amsterdam, October 2003
[9] M. Vilathgamuwa, J. Deng, K.J. Tseng, "EMI suppression with switching frequency
modulated DC-DC converters" Industry Applications Magazine-IEEE, Vol. 5, Issue 6,
pp 27 -33, December 1999
[10] D. M. Mitchell, "Power line filter design considerations for DC-DC converters",
Industry Applications Magazine-IEEE, Vol. 5, Issue 6, pp 16 -26, December 1999[11] Marian K. Kazimierczuck, Giusseppe Sancineto, Gabriele Grandi, Ugo Reggianiand Antonio Massarini, "High-Frequency small signal model of ferrite core inductors"
EEE Transactions on Magnetics, Vol. 35, N-5, pp 4185-4191, September 1999[12] W.D. Kimmel, "Wide frequency impedance modeling of EMI ferrites", Proc IEEE
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USA, pp 221 -223, August 1994122

3.7. References[13] Christopher Olivier, "A new core loss model" Switching power magazine, pp28-30, Spring 2002.
[14] M.J. Nave, "On modeling the common mode inductor", Proc. IEEE 1991
International Symposium on Electromagnetic Compatibility, 1991, Cherry Hill, NJ,
USA, pp 452-457, August 1991.
[15] F. Amemiya, K. Takagi, N. Kuwabara, S. Hamada, Y. Iwamoto, "Developing acommon-mode choke coil with high-permeability core used high-speed telecommunications
port for UTP cable", Proc. 2002 IEEE International Symposium on Electromagnetic
Compatibility Vol. 1, N-5, pp 314-319. September 2002.
[16] P. F. Okyere, "A novel physically-based Pspice compatible model for common mode
chokes", Proc. of International Symposium on Electromagnetic Compatibility, Tokyo,
Japan, pp.33-36, May 1999.
[17] Robert West , "Common mode inductors for EMI filter require careful attention to corematerial selection", PCIM magazine, July 1995.
[18] P. F. Okyere, L. Heinemann, "Computer-aided analysis and reduction of conducted
EMI in switched-mode power converter", Proc. Applied Power Electronics Conference
and Exposition, APEC '98, Anaheim, CA, USA, Vol. 2, pp 924-928, 1998.
[19] Michael Tao Zhang, "Electrical, thermal and EMI design of high density, low profile
power supplies", Dissertation, ETD-173510281975580, Virginia Polytechnic institute,
February 1997.
[20] D. Dallas, "Gain Confidence In Network Designs By Simulating The LAN
Magnetics", Electronic Design Magazine, Vol. 48, N- 2, pp 121-128 March 2000.
[21] Feng Zhang, "Investigation of electromagnetic interface of PWM motor drives in
automotive electrical systems", LEES Technical report TR-99-004, Massachusetts
Institute of Technology (MIT), December 1999.
[22] Keith Armstrong, "Choosing and using filters", EMC & Compliance Journal,
http://www.compliance-club.com/[23] Keith Armstrong, "Design Techniques part 3 - Filters and surge protection devices",
EMC & Compliance Journal , http://www.compliance-club.com/.
[24] Liu Jinjun, T.G. Wilson, T.G.; Zhao Qun, Dong Wei, F.C. Lee, "Large-signal modelof a downstream Dc/DC converter for analysis and design of front-end PFC rectifier using
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3. Conducted noise[25] B. Allonge, F. Arteche, C. Rivetta, F. Szoncso, "EMI Filter Design and Stability
Assessment of DC Voltage Distribution based on Switching Converters", Proc. 7thWorkshop on Electronics for LHC experiments, LECC 2001. Stockholm, pp 353-357,
September 2001.
[26] A. Ciccolella, T, Blancquaert, J. Verniolle, "Power inter-face characterization
between the Russian module and the ESA automated transfer vehicle", Proc. Sixth
European Space Power Conference, May 2002.
[27] C. Wildrick et.al., "A method of defining the load impedance specification for a Stable
Distributed Power System", IEEE Tranactions on Power Electronics, Vol.10, N-3, pp
280-285. May 1995
[28] Choi, B. Cho, B, "Intermediate Line Filter Design to Meet Both ImpedanceCompatibility and EMI Specifications", IEEE Transactions on Power Electronics,
Vol.10, N-5, pp 583-588, September 1995.
[29] L.R. Lewis,B.H. Cho, F.C. Lee, B.A. Carpenter, "Modeling, analysis and design ofdistributed power systems", Proc. Annual IEEE Power Electronics Specialists
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Trans. on Industry Applications, Vol. 25, N-6, 1989.
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124

Chapter 4
Noise Propagation

Power supply units for the CMS detector will be placed in two areas, the peripheryof the detector, distant about 20-40 meters from the front-end electronics and thecounting room located about 100-120 meters away from the FEE. The front-end
electronics and the power supply units are connected through long power cables
that propagate the output noise from the power supplies to the detector.
The understanding of the effect of long shielded power cables in the noisepropagation and its interaction with the FEE becomes an important issue to define
FEE immunity levels and the conducted noise emission levels required for the
power supplies. In addition, near and far electromagnetic fields induce currents inthe cable shields that interfere with the system if those currents are coupled to the
conductors feeding the FEE. This coupling depends on the surface transfer
impedance and surface transfer admittance of the shielded cable, which define thequality and cost of the cable.
In this chapter, the analysis of the conducted noise propagation through long cables
and its effect in the system are presented. The balance among cable parameters,
filters, and ground connection defines the effect on the FEE performance ofcommon mode (CM) and differential mode (DM) noise generated by the power
supplies. The influence of these parameters is studied using numerical simulation
based on multi-conductor transmission line models whose parameters are extracted
from real multi-conductor samples. The CM and DM propagation as well as the
cross-conversion between CM and DM conducted emissions are addressed in thisstudy.

4.1 Introduction to Multi-conductor
Transmission lines (MTL)
The traditional model of a multi-conductor transmission line is a distributed
parameter, multi conductor transmission line equation. There are a set of (n+1)

1254. Noise Propagation

partial differential equations representing a line consisting of n conductors and areference conductor for the line voltages. The line is considered to be a uniform linein that the (n+1) conductors are parallel to each other and the z-axis of a rectangularcoordinate system as it is shown in figure 4.1

Figure 4.1: MTL model.
Based on the previous picture, the system can be modeled by the set of partial
differential equations:∂∂
V ( z, t ) = – R ⋅ I ( z, t ) – L ⋅ I ( z, t )
∂z
∂t

(4.1)

∂
∂
I ( z, t ) = – G ⋅ V ( z, t ) – C ⋅ V ( z, t )
∂z
∂t

(4.2)

Where I(z,t) and V(z,t) are vectors representing the current and the voltage respect
to the reference conductor, respectively, L,C,R,G are the per unit length inductance,
capacitance, resistance and conductance NxN matrices representing thecharacteristic properties of the line and z is the position along the transmission line.
This model assumes transverse electromagnetic (TEM) waves as the propagation
mode. It allows a cable to be represented by per-unit parameters, which are
relatively easy to measure or calculate given the geometry of the cable.
A detailed analysis of MTL is presented in [1]-[4]. In the present chapter only thebasic concepts used to analyze the cables are defined. For the case of a genericmulti-conductor cable, figure 4.2 shows a scheme of an infinitesimally length dz ,where the parameters per-unit length have been identified. For the case of shieldedcables, the shield has been considered as the reference conductor.

1264.2. Definitions of per-unit length parameters

Figure 4.2: Distributed MTL model

4.2 Definitions of per-unit length parameters
Based on the TEM model of the transmission line, L,C,R,G parameters can becalculated as static field solution in the transversal plane for perfect line conductors.
The definitions of these matrices are given in the following paragraphs.

4.2.1 Per unit length inductance matrix
The per-unit length inductance matrix L, relates the current flowing in eachconductor with the flux linking the circuits formed by the conductors.

φ = L⋅I(4.3)

The diagonal elements represent the self-inductance, that are defined as the ratio
between the flux linking the loop formed by the i-th conductor and reference andthe current flowing through the i-th conductor.

φil ii = ---Ii(4.4)
I1 = … = Ii – 1 = Ii + 1 = … = I n = 0127

4. Noise Propagation

The mutual inductances l ij are defined as the ratio between the flux linking theloop formed by the i-th conductor and reference and the current flowing throughthe j-th conductor.

φil ij = ---Ij(4.5)
I 1 = … = I j – 1 = Ij + 1 = … = In = 0

l 11 l 12 … l 1n
L =l 21 l 22 … l 2n

… … … …
l n1 l n2 … l nn

(4.6)

The definition of the i-th circuit is critically important in obtaining the correct value
and the sign of these elements. The i-th circuit is the surface between the reference
conductor and the i-th conductor (which is of arbitrary shape but is uniform along
the line). The magnetic flux per-unit-length penetrating this surface is defined as
being in the clockwise direction around the i-th conductor when looking in the
direction of increasing z. In other words, the flux direction, Ψ i , through surface
s i , has the direction of the magnetic flux that would be generated by the currentof the i-th conductor. This is shown in figure 4.3.

Figure 4.3: Flux circuits for the estimation of the per-unit length inductances for a ribbon
cable structure of three conductors.

128

4.2. Definitions of per-unit length parameters

4.2.2 Per unit length capacitance matrix
The per-unit length capacitance matrix, C, relates the line voltage with the total
charge in each conductor.

Q = C⋅V(4.7)

The coefficients of the matrix C are the electrostatic induction coefficients.

Σ ( C 1k ) – C 12 … – C 1n
C =– C 21 Σ ( C 2k ) … – C 2n…
– C n1… … …
– C n2 … Σ ( C nk )(4.8)

The diagonal terms represent the self -electrostatic induction coefficients,n

Q i = V i ⋅ B ii = Vi ⋅ ∑ C ik

(4.9)

k=1

where Vj = 0 for all j ≠ iThese coefficients define a relation between the charges stored in the i-th conductor
with the voltage applied to the i-th conductor when the rest of conductors areconnected to the reference. Each self-induction coefficient is equal the sum of self
and mutual capacitances defined by the conductor.
The off-diagonal terms are the mutual-electrostatic induction coefficients anddefined as:

Q i = B ij ⋅ V i = – C ij ⋅ V i(4.10)

where Vi = 0 for all i ≠ jThis represents the charge stored in the i-th conductor when a unit voltage isapplied to the j-th conductor with all other conductors connected to the reference.
The elements of the C matrix are frequently referred to as the electrostatic induction
coefficients. However, these coefficients are not equal to the self and mutual
capacitance among the conductors and the term "electrostatic induction
coefficients" might be confusing. The self and mutual capacitance coefficients areshown in figure 4.4.

129

4. Noise Propagation

Vi − V j

j

i
C ij

Vj

C ii

C jj
0

Vi

Figure 4.4: Circuits of the definition of the per-unit length capacitance coefficients: self
terms and mutual terms.
A simple form to obtain the electrostatic inductance coefficients is by definition of
the potential matrix P.

V = P⋅Q(4.11)The entries in P are referred to as the coefficients of potential. Once the entries in [P]are obtained, the electrostatic induction coefficients are obtained taking the inverse–1of [P] ( [ C ] = [ P ] ) .

Vip ii = ----qiVil ij = ----Ij(4.12)
q q1 = … = q i – 1 = q i + 1 = … = q n = 0(4.13)
q1 = … = qj – 1 = qj + 1 = … = q n = 0

This method is usually used in finite element programs to calculate the capacitancematrix based on the geometry and material properties of the transmission line. In
the section Measurement Procedure1, a measurement method to determine both theself and mutual-capacitances and the electrostatic induction coefficients ispresented.

1 See section 4.3
1304.2. Definitions of per-unit length parameters

4.2.3 Per unit length resistance matrix
The per-unit length resistance matrix, R, relates the current flowing in eachconductor to the line voltage drop

R =R1 + Ro

Ro

…

Ro

Ro

R2 + Ro …

Ro

…
Ro

…
Ro

… …
… Rn + Ro(4.14)The diagonal elements of the matrix correspond to the sum of the per unit length
resistance of the i-th conductor and the resistance of the reference conductor. The off
diagonal of the matrix correspond only to the per-unit length resistance of thereference conductor.

4.2.4 Per unit length conductance matrix
The per unit length conductance matrix G, relates all the line voltages to the totaltransverse conduction current passing between the conductors.

It = G ⋅ V(4.15)

Σ ( g 1k ) – g 12 … – g 1n
G =– g 21 Σ ( g 2k ) … – g 2n…
– g n1… … …
– g n2 … Σ ( g nk )(4.16)The diagonal elements of the conductance matrix correspond to the sum of
self-conductance and mutual-conductance of conductor i, while the off diagonal
elements of the matrix corresponds to the negative value of the coefficient mutual
conductance between i-th and j-th conductor.
To determine the entry in G in the i-th row and j-th column we could enforce a
voltage between the j-th conductor and the reference conductor, with all otherconductor voltages set to zero and determine the transverse current, I ti , flowing
between the i-th conductor and reference conductor, as it is shown in figure 4.5

131

4. Noise Propagation

Figure 4.5: Circuits of the definition of the per-unit length conductance coefficients.
Denoting each of the entries in G as

I ti
G ij = ----Vj

G ij

we have:(4.17)

V q1 = … = V i – 1 = V i + 1 = … = V n = 0

In general the values of this matrix are very small in the range of frequencies of ourinterest and G-matrix can be considered as a null matrix. For this reason thesevalues have not been measured.

4.3 Parameters measurement procedure
All per unit values defined above can be measured or calculated based on thegeometry and material properties. In the present study some of them are calculated
based on [4] and some of them are measured using an LRC meter covering afrequency range between 1 kHz and 5 MHz. Some of these parameters are
frequency dependent due to skin1 effect in the conductors and proximity effects.
The measurement procedure is presented and results from HCAL power cable andTracker power cable are used as an example of these measurement procedure.

4.3.1 InductanceLet us consider a tracker power cable as a multi-conductor cable with three
conductors and a reference line (for the HCAL power cable this is actually a two
conductor cable). In both cases, the shield of the cable is considered as the reference
1 A detailed analysis of this effect is presented in appendix A.
132

4.3. Parameters measurement procedure

conductor. The aim of the measurement procedure is to calculate the 6 values (3 for
HCAL) of the multi-conductor theory inductance matrix, which is symmetrical. The
self-inductance ( L 11, L 22, L 33 ) and mutual-inductance between conductors( L 12, L 13, L 23 ) are measured to complete the inductance matrix.
The dependence of the cable inductance with the frequency and the cableresonances force the use of cables with different length to carry out the measure of
the cable inductance in the whole frequency range. Short cables are used to performthe measurement of the inductance at high frequency to avoid the resonance ofcable and to have completely developed the skin effect. However long cables areused to measure the inductance at low frequency, where the skin effect is not stilldeveloped. At low frequency, the cable length has to be compatible with minimum
impedance and precision that the instrument can measure.
The values of the self-inductance of the cable have been measured with thefollowing configuration. The line to be measured and the shield are short-circuited.
All remaining lines are left in an open circuit configuration. Figure 4.6 shows thelayout to measure the inductance L ii . The measured value is the inductance
between the i-th and reference conductor.

L13

L12

L23

Figure 4.6: Cable connection to calculate the Self inductance of line 2.5V of Tracker power
cable (MSC).
The values of each mutual inductance of cable can be measured with the following
configuration shown in figure 4.7. The lines, whose mutual inductance will bemeasured, should be connected together and all remaining lines and the reference
are left in an open circuit configuration.

133

4. Noise Propagation

L13

L12

L23

Figure 4.7: Cable connection to calculate the Mutual inductance of line 1 and 2 for Tracker
power cable (MSC).

L ii + L jj – M ij
L ij = --------------------------------2(4.18)

Where the M ij is the value measured with the LRC meter. The same procedure isused to obtain the rest of the values of the matrix. The measured values for bothcable samples are shown in figure 4.8 and 4.9.

Figure 4.8: Measured values of per-unit length inductance matrix for Tracker power cable(MSC).

1344.3. Parameters measurement procedure

Figure 4.9: Measured values of per-unit length inductance matrix for the HCAL power
cable.
From the above picture it is possible to see that the inductance values are dependentof the frequency due to skin effect. The total per unit length inductance of atransmission line is the sum of the external and internal inductance. The external
inductance is frequency independent and depends mainly of the external
configuration of the cable. However, the internal inductance is approximately
constant (for a single wire is equal to 50 nh/m) only in the low frequency range, up
to a frequency where the conductor radius is on the order of skin depth. Above thisfrequency the conductor internal inductance decreases with an order of f or 10 dB
per decade. Thus the total inductance at low frequencies is equal to:

l = li + le(4.19)

At high frequencies the internal inductance can be neglected and the total
inductance can be approximated by:l ≈ le(4.20)

4.3.2 CapacitanceLet us consider the three-conductor cable of the tracker power cable(Two-conductors for HCAL) and a reference line. The self and mutual -capacitanceare defined as depicted in figure 4.10.

135

4. Noise Propagation

Figure 4.10: Self and mutual capacitance of Tracker power cable (MSC).
To identify the capacitances shown in figure 4.10 one performs a series ofmeasurements, that are the dual with respect to the ones performed to obtain the
inductances. This technique is well described in [5] and [6], hence in this section
only two particular cases are analyzed. Let us denote Cm ij as the measured value,that is, the capacitance between the i-th and j-th conductors following theconnections indicated in figure 4.11.

Cd 1

Cd 2

Cd 3

Cd 1

Cd 2

Cd 3

Cd 3

Ref
Cd 1

Cm12

Cd 2

Cd 3

Ref
Cd 1

Cd 2

Cd 3

Cm13

Cm 23
Ref

Cd 3

Cm33

Ref
Cd 2

Cd 2

Cm22

Cm11
Cd 1

Cd 1

Ref

Ref

Figure 4.11: Cable connection to calculate the capacitance matrix values of Tracker power
cable (MSC).
The lines in this figure indicate a short-circuit connection between the conductor
and ground or between the conductors. The definition of Cm ij and C ij involves
2(n – n)2symmetry relations and we need to establish ------------------- + n relations instead of n2
(6 unknowns values for the tracker instead of 9, and 3 unknowns instead of 4 for the
HCAL power cable).

1364.3. Parameters measurement procedure

Based on the definition given in figures 4.10 and 4.11, it is possible to establish thefollowing relations.

Cm 11 = C 10 + C 12 + C 13

(4.21)

Cm 12 = C 10 + C 13 + C 22 + C 23

(4.22)

Cm13 = C 10 + C 20 + C 30

(4.23)

Cm 22 = C 12 + C 22 + C 23

(4.24)

Cm 23 = C 12 + C 22 + C 33 + C 13

(4.25)Cm 33 = C 13 + C 23 + C 33

(4.26)

To identify the capacitance coefficients and the electrostatic induction coefficients
based on the measurement of the equivalent capacitances Cm ij ,three different
vectors are defined by:t

Cm = [ Cm 11, Cm 12, Cm 13, Cm 22, Cm 23, Cm 33 ] - Measured vectort

C p = [ C 10, C 12, C 13, C 20, C 23, C 30 ] - Self / Mutual capacitance vector(4.27)(4.28)t

C = [ B11, B 12, B 13, B 22, B 23, B 33 ] - Electrostatic induction coefficient1 vector (4.29)
Based on equation 4.8 and 4.21 to 4.25, the vectors are related by the following
equations.

C = D

–1

⋅H

–1

⋅ Cm

Where the definitions of H and D–1

1 1 1 00 01 0 1 11 0
H = 1 0 0 10 10 1 0 11 0
0 1 1 10 1
0 0 1 01 1

; Cp = H–1⋅ Cm(4.30)are:1 1 1 0 0 0
0 –1 0 0 0 0–1
D = 0 0 –1 0 0 0
0 1 0 1 1 0
0 0 0 0 –1 0
0 0 1 0 1 1(4.31)

1 Multi conductor theory capacitance matrix coefficients are defined by equation
4.8

137

4. Noise Propagation

The measured values are shown in figure 4.12 for tracker (MSC) and 4.13 for the
HCAL power cable. As it can be observed from the plots, the magnitude of thecapacitance is not as strongly frequency dependent [7] as the inductance.

Figure 4.12: Measured values of per-unit length Cm-vector for the Tracker power cable(MSC).

Figure 4.13: Measured values of per-unit length Cm-vector for the HCAL power cable

4.3.3 ResistanceThe goal of this measurement is to calculate the 4 values of the multi-conductor
theory resistance matrix for the tracker (3 values for HCAL power cable). The
resistance of each conductor ( R 1, R2, R 3 ) and the resistance of the shield ( R 0 ) areneeded to complete the resistance matrix. These values were measured at the same138

4.3. Parameters measurement procedure

time as the inductance was measured with the LRC meter. The measured values are( R 11, R22, R33, R12, R23, R13 ). The values of the diagonal can be obtained directly andthe value of R 0 is easily calculated by solving an equation system. From theconductor and shield resistance values, the resistance matrix (equation 4.14) can becalculated. Figure 4.14 and 4.15 show the values of the resistance per unit length asa function of the frequency for Tracker power cable and HCAL power cable.

Figure 4.14: Measured values of per-unit length resistance for the Tracker power cable(MSC).

Figure 4.15: Measured values of per-unit length resistance for the HCAL power cable.
1394. Noise Propagation

4.4 Frequency domain solution of MTL systems.
Transmission lines are distributed parameter systems. If the electrical dimensions ofa structure are small in comparison with the wavelength of the signal considered,
the line can be modeled as a lumped-parameter system. The independent variables
for a distributed parameter system are the spatial dimension, x, y, z, and time. In thecase of a lumped parameter system, the quantities of interest are lumped rather
than distributed throughout the space so that they depend only on time. Lumped
parameters systems are characterized by ordinary differential equations (ODE),
whereas distributed-parameters systems, such as transmission lines, arecharacterized by partial differential equations.

∂∂
V ( z, t ) = – R ⋅ I ( z, t ) – L ⋅ I ( z, t )∂z∂t(4.32)∂∂I ( z, t ) = – G ⋅ V ( z, t ) – C ⋅ V ( z, t )∂z∂t(4.33)As the frequency domain solution is considered, the use of phasor quantities
removes the time dependence. In the case of transmission lines the only spatial
parameter is the line axis, z, along the transmission line and, in frequency-domain,
the partial differential equations become ordinary differential equations withindependent variable z and complex value coefficients. This simplification issummarized in equations 4.34-4.35.

d ˆ
V ( z ) = – Zˆ ⋅ ˆI ( z )dz(4.34)dˆI ( z ) = – Yˆ ⋅ Vˆ ( z )dz(4.35)

Assuming G=0, the impedance and admittance per unit length are defined by:Zˆ = R + j ⋅ ω ⋅ L(4.36)

Yˆ = j ⋅ ω ⋅ C(4.37)j⋅ ω ⋅ twhere Vˆ ( z ) and ˆI ( z ) are phasors defined as Vˆ ( z ) = V ( z ) ⋅ e. where j = – 1 ,ω = 2 ⋅ π ⋅ f and L,R,C are the parameter matrices defined in the previous
sections.

Equations 4.34 and 4.35 may be presented in a more compact form.

d ˆX ( z ) = Aˆ ⋅ Xˆ ( z )dz
140(4.38)

4.4. Frequency domain solution of MTL systems.

ˆ (z) =
Where XVˆ ( z ) andˆI ( z )Aˆ ( z ) =0 – Zˆ
– Yˆ 0(4.39)

Thus, it is possible to make a direct analogy between the sinusoidal steady statetransmission-lines equation and those of general lumped-parameter systemrepresented by an ODE, by viewing the spatial parameter, z, in the
distributed-parameter system phasor equations as the equivalent of time, t, in the
lumped-parameter system governing equations.
Based on this analogy, which is well explained in [3] [4], the general solution for thelines voltages and currents is:Vˆ ( L ) = Φ̂ 11 ( L ) Φ̂ 12 ( L ) ⋅ Vˆ ( 0 )ˆI ( L )ˆI ( 0 )Φ̂ 21 ( L ) Φ̂ 22 ( L )(4.40)

Where 0 represents the beginning of the line and L the end point of the transmission
line.
The matrix Φ̂ ( L ) is the NxN state-transition matrix and it has several properties as:Φ̂ ( 0 ) = I 2n ,(4.41)

where I 2n is the unit NxN matrix.
–1Φ̂ ( L ) = Φ̂ ( L )(4.42)

2

2
LLΦ̂ ( L ) = I 2n + ----- ⋅ Aˆ + ----- ⋅ Aˆ + …
1!
2!

(4.43)

So the essential task in solving the MTL equations is to determinate the chain
parameter matrix Φ̂ ( L ) . This is generally done by the modal method. The idea is todefine the characteristic mode, which propagates in the multi-conductor line. In thiscase, there will be different natural propagations modes and also different
characteristic impedances. These characteristics modes are calculated transforming
the propagation equations into an uncoupled model, which can be easily solved.
This transformation is achieved by obtaining the second order differential equationsof 4.34 and 4.35:

d⎛d ˆ ⎞dV ( z ) = – Zˆ ⋅ Iˆ ( z )⎠dz⎝ dzdz(4.44)

141

4. Noise Propagationd⎛dˆ ⎞dI ( z ) = – Yˆ ⋅ Vˆ ( z )⎠d z⎝ d z
dz(4.45)

Then, the first derivates of current and voltage of equations 4.44 and 4.45 aresubstituted by the values of equation 4.34 and 4.35, which leads to:2d ˆ
V ( z ) = Zˆ ⋅ Yˆ ⋅ Vˆ ( z )2dz(4.46)2d ˆI ( z ) = Yˆ ⋅ Zˆ ⋅ ˆI ( z )2dz(4.47)

It is possible to find a similarity transformation Tˆ v and Tˆ i , which simultaneously
diagonalize the per-unit length parameter matrices, Zˆ , Yˆ , Zˆ ⋅ Yˆ and Yˆ ⋅ Zˆ . The
transformation essentially changes a n-coupled, first order differential equation set
to an n-uncoupled differential equation set. This similarity transformation
preserves the eigen-values of the original matrices, which are known as the
propagation constants.
The matrices that achieve such transformation are:Tˆ v - is the matrix whose columns are the complex eigen-vectors of Zˆ ⋅ Yˆ .
Tˆ i -is the matrix whose columns are the complex eigen-vectors of Yˆ ⋅ Zˆ .
These matrices have the following property:–1ˆtTi = Tˆ v(4.48)

Applying the transformation, the equations of the modal system are obtained(Eq.4.49 and 4.50):2d ˆˆ –1 ˆ ˆ ˆ ˆVm(z ) = T v ⋅ Z ⋅ Y ⋅ T v ⋅ V m( z)
2
dz
2
–1
d ˆ
I m ( z ) = Tˆ i ⋅ Yˆ ⋅ Zˆ ⋅ Tˆ i ⋅ Iˆ m ( z )2dz(4.49)

2

2 ˆ
d ˆ
V
m ( z ) = γˆ ⋅ V m ( z )
2
dz
2
2
d ˆ
I m ( z ) = γˆ ⋅ Iˆ m ( z )
2
dz

142

(4.50)

4.4. Frequency domain solution of MTL systems.

2
γˆ
1

0 … 0

0 γˆ

2
γˆ =

2

… 0

2

(4.51)… … … …

0

0 … γˆ

2n–1–1where Vˆ m ( z ) = Tˆ v ⋅ Vˆ ( z ) and the ˆI m ( z ) = Tˆ i ⋅ ˆI ( z ) . Each value of the diagonalcorresponds to the propagation constant of the i-th mode. γˆ is the matrix
containing the complex eigen-values of, Zˆ ⋅ Yˆ and Yˆ ⋅ Zˆ .

The general solutions of these uncoupled equations are:

ˆI ( z ) = e –γzˆ ⋅ Iˆ + – e γzˆ ⋅ Iˆ m
m
m

(4.52)

ˆ
ˆ
ˆ+
ˆ– γz
γz
Vˆ m ( z ) = e ⋅ V m + e ⋅ Vm

(4.53)± ( γ̂1 ⋅ z )ˆ± γzwhere e

=

0
…

…

0

…

…
…

0
…

0

… e

0

e

± ( γ̂2 ⋅ z )e

0(4.54)± ( ˆγn ⋅ z )The actual voltages and currents can be obtained by multiplying this result by thetransformation:
–1 t
ˆ
ˆ
ˆ+
ˆ– γz
γz
Vˆ ( z ) = ( Tˆ ) ⋅ ( e ⋅ V m + e ⋅ Vm )(4.55)

ˆI ( z ) = Tˆ ⋅ ( e – γzˆ ⋅ Iˆ + – e γzˆ ⋅ Iˆ - )m
m(4.56)Tˆ = Tˆ i(4.57)

Evaluating previous equations, 4.55 and 4.56, at both ends of the lines ( z = 0 andz = L ) and substituting the results into equation 4.40, the components of the chain
parameter matrix may be evaluated.
–1
ˆ
ˆ
1 –1
γL
– γL
Φˆ11 ( L ) = --- ⋅ Yˆ ⋅ Tˆ ⋅ ( e + e ) ⋅ Tˆ ⋅ Yˆ2

(4.58)

143

4. Noise Propagation

–1
ˆ
ˆ
1 –1
γL
– γL
Φˆ12 ( L ) = – --- ⋅ Yˆ ⋅ Tˆ ⋅ γˆ ⋅ ( e – e ) ⋅ Tˆ2

(4.59)

–1
ˆ
ˆ
–1
1
γL
– γL
Φˆ21 ( L ) = – --- ⋅ Tˆ ⋅ ( e – e ) ⋅ γˆ ⋅ Tˆ ⋅ Yˆ2

(4.60)

–1
ˆ
ˆ
1
γL
– γL
Φˆ22 ( L ) = --- ⋅ Tˆ ⋅ ( e + e ) ⋅ Tˆ2

(4.61)

At each frequency the matrix elements Φ̂ ij ( L ) are constant and confirm theproperties shown in equation 4.41, 4.42 and 4.43.

4.4.1

Solution of MTL incorporating the terminal conditions.

The chain parameter matrix given in 4.38 relates the phasor voltage at z=0 and at
z=L, but does not explicitly determine these voltages and currents. Essentially, 2nrelations are needed to explicitly determinate the terminal voltages and currents
from the chain parameter matrix relation. These relations are provided by theterminal constraints. The aim of this section is to incorporate these terminal
constraints to explicitly determine the terminal voltages and currents and complete
this important last step in the solution.
There are many ways of relating the voltages and currents at the terminals of an n
port network. If the network is linear, this relationship will be a linear combinationof the port voltages and currents. One obvious way is to generalize the Thevenin
equivalent representation of a 1 port as:Vˆ ( 0 ) = VˆS + ZˆS ⋅ ˆI ( 0 )(4.62)Vˆ ( L ) = VˆL + Zˆ L ⋅ ˆI ( L )(4.63)The Nx1 vectors VˆS and Vˆ L contain the effects of the independent voltage andcurrent in the termination networks at z=0 and z=L, respectively. The NxNmatrices, ZˆS and Zˆ L contain the effects of the impedances and controlled sourcesin the terminal networks at z=0 and z=L, respectively.
In general, the impedance matrices ZˆS or Zˆ L , have diagonal and off diagonalcomponents. However, there may be terminal-network configurations where these
impedances matrices are diagonal and the only coupling occurs along the MTL.

144

4.4. Frequency domain solution of MTL systems.

Figure 4.16: The Generalized Thevenin representation at source level
Figure 4.16, shows such a case where in each line at z=0 is terminated directly to thechosen reference conductor with an impedance and a voltage source. In this case thematrices become:Vˆ S =Vˆ S 1
Vˆ S
2

…
Vˆ S

n

Vˆ L =Vˆ L 1
Vˆ L
2

…
Vˆ L

n

Zˆ S 1 0 … 0ˆZˆS = 0 Z S2 … 0
0 0 … 00 0 … Zˆ S

(4.64)Zˆ L 1 0 … 0ˆZˆ L = 0 Z L2 … 0
0 0 … 00 0 … Zˆ L

(4.65)nn

The generalized Thevenin equivalent characterizations in equation 4.62 and 4.63 aresubstituted into the chain parameter matrix characterization given in 4.40, to solve
the MTL equations with the terminal conditions.

Vˆ ( L ) = Φ̂ 11 ( L ) ⋅ Vˆ ( 0 ) + Φ̂ 12 ( L ) ⋅ ˆI ( 0 )(4.66)

ˆI ( L ) = Φ̂ 21 ( L ) ⋅ Vˆ ( 0 ) + Φ̂ 22 ( L ) ⋅ ˆI ( 0 )(4.67)

To yield145

4. Noise Propagation( Φ̂ 12 – ZˆS ⋅ Φ̂ 11 – Zˆ L ⋅ Φ̂ 22 + Zˆ L ⋅ Φ̂ 21 ⋅ ZˆS ) ⋅ ˆI ( 0 ) = Vˆ L – ( ( Φ̂ 11 – ( Zˆ L ⋅ Φ̂ 21 ) ) ⋅ Vˆ S )(4.68)

ˆI ( L ) = Φ̂ 21 ⋅ Vˆ S + ( ( Φ̂ 22 – ( Φ̂ 21 ⋅ Zˆ ) ) ⋅ ˆI ( 0 ) )S(4.69)

Equation 4.68 is a set of n equations that defines the current at z=0 as function of theexternal voltages. Once these equations are solved, the n terminal currents at z=L,
ˆI ( L ) , can be obtained from 4.69. The n terminal voltages, Vˆ ( L ) and Vˆ ( 0 ) can beobtained from equation 4.62 and 4.63.
All of the equations presented above were incorporated in a MATLAB program to
study the signal propagation in power cables and also to analyze the susceptibilityof some connections of the HCAL-FEE. Other commercial software oriented tosolve electrical circuits as Pspice, Saber, etc. incorporate models to analyzetransmission lines. However the necessity of incorporating the skin effects, shields
currents and external fields perturbations, which will be analyzed later, forced us towork directly with the equations in MATLAB1.
There are many more studies and equivalent circuits that could incorporate theterminal conditions to the MTL equations such as Norton equivalent circuit and amixed representation of both Thevenin and Norton circuits. These representations
were not used in the present study and thus they are not explained here in detail. A
detailed bibliography of these configurations is shown in [2][4].

4.5 Noise propagation in cables
4.5.1 Introduction
The effects that electrically-long power cables have on the noise propagation havebeen addressed recently in the literature [8][9][10]. Some of those studies arefocused on predicting the effects introduced on the measurement of the conductive
noise, based on the EU standard, when the equipment under test is connected to thenormalized impedance using a long cable. In the present analysis, the shielded
twisted power cable of the HCAL sub-system has been chosen to study the effect ofnoise propagation in long power cables. In this part of the analysis, it is assumed
that no current flows through the cable shield. The study is focused on the
input/output transfer functions of a 15 mts. shielded twisted power cable fordifferent load conditions. The topology of the circuit under study is shown in figure
4.17.

1 Program validation in appendix A
146

4.5. Noise propagation in cables

V dm , Vcm ( 0 ) & I dm , I cm ( 0 )Cable Vdm , Vcm ( L) & I dm , I cm ( L)

Noise sources

VDM
VCM

Z L2

ZL1
Load

Figure 4.17: Circuit layout for the noise propagation analysis.
The per unit length parameter matrices for this cable are taken from themeasurements of the HCAL power cable presented in section 4.3. The chosen values
are measured at 1-MHz. The values of the inductance and capacitance per unit
length matrix are:
–6 H
L = 0.1479 0.0260 × 10 ---m
0.0260 0.1540

C =(4.70)

F
0.2292 – 0.0412 × 10– 9 --m– 0.0412 0.2339(4.71)

The cable model includes the skin effect in the central conductors and in the shield.
According to the skin effect modelization presented in [4], the frequency
dependence of the resistance of a conductor may be approximated as:f
R ( f ) = R DC + R DC ⋅ ⎛ ---- ⎞⎝ fo ⎠(4.72)The value of f o is the frequency where the skin depth is half of the conductor
radius. This approximation has a deviation at the transition frequency but for therequired frequency domain analysis, the error is minimized. The value of theresistance for the HCAL power cable matrix is found to be:

R =

f
f
0.0017 ⋅ ⎛ 1 + ------------⎞ + 0.0035 ⋅ ⎛ 1 + ------------------⎞
⎝
⎝
5400⎠
160000⎠f ⎞⎞⎛ 0.0035 ⋅ ⎛ 1 + ----------------⎝⎝
160000⎠ ⎠f ⎞⎞⎛ 0.0035 ⋅ ⎛ 1 + ----------------⎝⎝
160000⎠ ⎠ff
0.0018 ⋅ ⎛ 1 + ------------⎞ + 0.0035 ⋅ ⎛ 1 + ------------------⎞
⎝
⎝
5400⎠
160000⎠Ω---m(4.73)

Four load configurations have been chosen to study their influence in the noisepropagation:

147

4. Noise Propagationa)( Z L1 = Z L2 = Z c )b)( Z L1 = Z L2 = 150 Ω )c)

Z L1 = 150 Ωd)

Z L1 = 150 Ω C = 1 µ f

Z L2 = 0 Ω
Z L2 = 0 Ω

For the first configuration, the load impedances are equal to the characteristic
impedance of the cable. For the analysis of the CM noise propagation, they are set
equal to Z c – cm = 30.4 Ω and for the DM noise propagation, they are equal toZ c – dm = 21.4 Ω . This case is unpractical for power distribution and it is included in
the study as reference. For the second one, a balanced impedance of 150 Ω isselected as load. This case is chosen because 150 Ω is used as the normalized
impedance for studies in CM propagation in cables. Finally the last two
configurations represent unbalanced loads. In this condition, one of the lines is
directly connected to ground and the other is loaded with a resistor larger than thecharacteristic impedance of the cable. The case d) includes a capacitor in parallelwith the 150 Ω resistor to simulate the effect of an input filter at the load-end.
The terminal configurations of the CM and DM circuit presented are characterized
as a generalized Thevenin equivalent as in (4.62 and 4.63) where:

ZL =Z L1 0Ω , V L = 0 V Load level (Both configurations)
00 Z L2Z S = 0 0 Ω , V SCM = 1 V and
0 0
1

V SDM =1 V
–1Source level(4.74)(4.75)

This study is conducted decomposing the voltage and current for each conductor, ateach position, in two orthogonal components. For currents, the common anddifferential mode are defined as:

148

I 1 ( z ) + I2 ( z )I cm ( z ) = -----------------------------2(4.76)I 1 ( z ) – I2 ( z )I dm ( z ) = -----------------------------2(4.77)

4.5. Noise propagation in cables
where I 1 ( z ) and I 2 ( z ) are the current flowing through conductor 1 and 2
respectively at the z position in the cable. Similar definition is used for voltages.

4.5.2 Common mode noise effects
To study the propagation of the common mode signals, based on the circuit
depicted in figure 4.18, the CM current transfer function is defined as:

Icm ( L )Tcm I = -----------------Icm ( 0 )(4.78)

where I cm ( 0 ) and I cm ( L ) are the CM current at the sending-end andreceiving-end of the cable. Figure 4.18 shows the simulated results of the CM
current transfer function for the four load configurations.

Figure 4.18: CM current transfer function - Simulated values.
For load resistors equal to the CM characteristic impedance of cable, the behavior issimilar to a low-pass filter where the attenuation is defined by the series resistanceof the conductors. When the load is not matched, cable resonances are leading in the
propagation of noise along the cable. For the balanced case and high impedanceload, resonances are present, however the CM current is attenuated when it
propagates along the cable. On the other hand, when the load impedance is149

4. Noise Propagation

balanced and lower than the characteristic impedance, the cable starts amplifying
the CM currents at the resonance frequencies. Unbalanced load displays similar
results to the balanced load condition. In the case d), where a filter capacitor isadded to an unbalanced load, both loads are connected to ground through low
impedances, the CM currents can be amplified up to 26 dB (20 times) at certain
frequencies. This situation might happen if a RF capacitor is placed at the load inorder to filter the noise of the line. This capacitor at high frequency presents very
low impedance to ground at the end of the line. This effect is important to take into
account because the CM current flowing through the cables defines mainly theradiated emission of the cables.
The common mode voltage at the load-end is defined by the CM current and theload impedance. For case d) the CM voltage at the load is effectively attenuated by
the load impedance while for the other cases, the CM voltage at the load is similaror larger than the CM mode voltage applied at the sending-end. To show thisbehaviour, the CM voltage transfer function, is defined as and plotted in figure 4.19.

Vcm ( L )Tcm V = ------------------Vcm ( 0 )(4.79)

Figure 4.19: CM voltage transfer function - Simulated values.
Another important aspect in this analysis is the CM to DM conversion. This
conversion is due to the load unbalance and also to the intrinsic minor unbalance ofthe cable. Figure 4.20 shows the CM to DM current transfer function for the four
configurations. This transfer function is defined as:

150

4.5. Noise propagation in cables

Idm ( L )Tcm – dm I = -----------------Icm ( 0 )(4.80)

Figure 4.20: CM to DM current transfer function - Simulated values.
For balanced loads (cases a) and b)), the CM to DM conversion exists due to theunbalances in the cable parameters, which are dominated by the mismatch of thecable inductance. From figure 4.20, it is possible to observe that this effect isimportant only at very high frequencies. For the case of unbalanced loads, theconversion is determined by the load and at some resonant frequencies theconversion mode can have a gain of 10 dB or more.
Figure 4.21 shows the CM to DM voltage transfer function for the four
configurations. This transfer function is defined as:

Vdm ( L )Tcm – dm V = -------------------Vcm ( 0 )(4.81)

151

4. Noise Propagation

Figure 4.21: CM to DM voltage transfer function - Simulated values.
This analysis is used to get a better understanding of the effect of unbalances of the
input power circuit of the FEE. The worst case is found when the system isunbalanced at load level, Z L1 = 150 Ω - Z L2 = 0 Ω -(in red). For the balanced-loadcase the dominant effect is the unbalance of the cable, which is basically defined by
the inductance of cable in the frequency range of analysis.

4.5.3 Differential mode noise effects
The DM current and DM current transfer functions of the circuit are defined as:

Idm ( L )Tdm I = -----------------Idm ( 0 )(4.82)

Figure 4.22 shows the DM current transfer function for the four load configurations
mentioned before.

1524.5. Noise propagation in cables

Figure 4.22: DM current transfer function - Simulated values.
Again, the resonant frequencies are only attenuated for values equal to the DM
characteristic impedance of cable Z L1 = Z L2 = 21.4 Ω , being amplified for low
impedances values. These results are very similar to the ones for the CM current
transfer function. However, their implications are not as important as in the CM
currents because the DM current ability to radiate is much lower than the CM
currents and also the FEE is more sensitive to CM perturbations flowing throughthe power cables than DM perturbations [11].
Plots 4.23, 4.24 and 4.25 show the DM voltage, DM to CM current and DM to CM
voltage transfer functions. The curves are very similar to the ones presented in the
CM analysis with only small differences in amplitude and resonant frequencies.
Basically both configurations are the same with the only difference being theimpedances of both CM and DM circuits.

153

4. Noise Propagation

Figure 4.23: DM voltage transfer function - Simulated values.
It is important to remark that the transfer functions (current and voltage) of CM to
DM and DM to CM conversion are very similar, confirming the close relation
between both modes.

Figure 4.24: DM to CM current transfer function - Simulated values.

1544.5. Noise propagation in cables

Figure 4.25: DM to CM voltage transfer function - Simulated values.

4.5.4 Summary of noise propagation
The study presented above considers different impedances at the load-end of thecable to demonstrate that both CM/DM currents and voltages can be amplified by
the cable resonance when they propagate from the sending-end to thereceiving-end. The only case where resonances do not exist is when the load
impedance matches the characteristic impedance of the cable, being for power
distribution completely unpractical.
High impedance, balanced loads attenuate the CM and DM currents transmitted
through the cable, but the CM and DM voltage across the load terminals is
amplified. On the other hand, low impedance and balanced loads amplify the CM
and DM currents at the resonance frequencies.
In high-energy physics applications, generally, the return plane of the FEE isconnected to ground to minimize the voltage variations across the parasitic
capacitances between the shielding box and the sensitive inputs of the amplifiers or
to provide a return to faults currents1. This connection implies in general, that thereturn power cable is connected to ground. From this study it is shown that for
unbalanced loads, both the CM and DM mode signal are amplified at the resonant
frequencies of the cable. In particular, when a capacitor filter is included across the
1 See chapter 2

155

4. Noise Propagationload terminals, the CM and DM currents flowing through the cable are amplified,
but the DM and CM voltage across the load terminals are effectively attenuated.
This configuration reduces the noise current coupled to the resistive part of the load
by-passing those currents through the capacitor. This behavior explains a typical
EMC-based design rule, which is to put the by-pass filter capacitor of the input
power filter as close as possible to the entrance point of the power cable [3]. Thisfilter limits both the DM and CM voltage at the input but amplifies both the CM andDM currents flowing through the cables. If the filter is placed close to the sensitiveelectronics into the screened box or cabinet, the power cable running inside the box,
from the entrance point to the filter, radiates and could interfere with the FEE.
The current levels produced by the amplification of the CM and DM currents for thecase of unbalanced loads with capacitor filter can generate unwanted
electromagnetic radiation from the cable. The radiation efficiency of a cable is
proportional to the loop area of the current path. For differential mode currents, theeffective area is minimum if the both conductors, active and return, are adjacent toeach other or are twisted. For CM currents, the return path is not always perfectly
defined. In the present analysis, it is assumed the CM current return through theinternal part of the shield and, in this case, the effective area is ideally zero. If thecable has no shield, the return path is not completely defined by the cable itself but
it is also defined by the position of the cable with respect to the reference structureand hence the effective radiation area can be large. It is critical for cables without
shielding to have reduced levels of CM current to avoid unwanted electromagnetic
radiation.

4.5.5 CM and DM mode noise propagation in HCAL power supply
distribution system
4.5.5.1 Introduction
In the previous sections, the effect that the cable and the load have on the conducted
noise propagation has been presented. In this section, the noise propagationinduced by power supplies (PS) and the noise generated by ground currents arestudied for the power supply distribution of the HCAL sub-system. In thissub-system, the nominal low voltages at the FEE input power are 6.5V and 4.5V andinternal linear regulators convert those values to 5V, 3.3V and 2.5V. Shielded
twisted-pair cables, whose length is between 15 and 20 meters, connect the power
supply units and the FEE.
In this analysis, the model is composed by three blocks: the PS unit and EMI filter,the cable and the FEE, as it is depicted in figure 4.26.

156

4.5. Noise propagation in cables

Vdm ,Vcm (0) & I dm , I cm (0)Vdm ,Vcm ( L) & I dm , I cm ( L)

ICM

Cable
EMI Filter
VDM

IDM

FEE
ICM

PS UnitFigure 4.26: HCAL power supply distribution layout
The PS unit and EMI filter are represented by the combination of two current
generators (CM and DM) and output capacitors. These sources represent theemissions generated by a DC-DC converter. The EMI filter is represented by threecapacitors, one DM capacitor of 1
and two CM capacitors of 5 nF . Powersupplies are connected to the FEE through a 15 meter-long cable1 (external diameterof 12 mm), placed 10 mm above a metallic tray. The FEE input power is represented
by a reduced model composed by a set of real capacitors of 1 µ F , with parasitic
resistance is equal to 0.2 Ω , in parallel with a 1 M Ω resistance.
4.5.5.2 Common mode noise propagation
The terminal configurations of the CM circuit are characterized by a generalized
Thevenin equivalent as in (4.62 and 4.63) where:

ZS =Z cm1 ⋅ ( Z cm2 + Z dm )Z cm1 ⋅ Z cm2------------------------------------------------ --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Zdm
Z cm1 ⋅ Z cm2
Z cm2 ⋅ ( Z cm1 + Z dm )--------------------------------------------- -----------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + ZdmΩ(4.83)

1
1
1
Z cm1 = ------------------------ , Z cm2 = ------------------------ and Z dm = -----------------------j ⋅ ω ⋅ C cm
j ⋅ ω ⋅ C cm
j ⋅ ω ⋅ C dm
correspond to the impedance of the output filter of the power supply.
The values of

1 The per-unit length matrices of the cable are presented in equations 4.70 and
4.71.

157

4. Noise Propagation

VS =

V S1

=

V S2

( Z cm2 + Z dm )Z cm1--------------------------------------------- --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dm
Z cm2
( Z cm1 + Z dm )--------------------------------------------- --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dm⋅

I cm1 ⋅ Z cm1
I cm2 ⋅ Z cm2

Volts(4.84)

Where the I cm1 = 1 and I cm2 = 1 .

ZL =Where Z L2 = 0 Ω

Z L1 0Ω , V L = 0 V Load level
00 Z L2(4.85)

Z L1 = 150 Ω C = 1 µ f + 0.2 Ω.

Figure 4.27 shows the CM current and voltage transfer function of the system
between the output of the filter and at the input of the FEE.

I cm ( L )V cm ( L )TFcm – I = ---------------- and TF cm – V = ----------------I cm ( 0 )V cm ( 0 )(4.86)

Figure 4.27: CM current and voltage transfer function of HCAL power supply distribution
system - Simulated values.

1584.5. Noise propagation in cables

The low impedance presented by the FEE at high frequency, amplifies the CM
currents at the resonance frequencies but effectively attenuates the CM voltage at
the input terminals. Based on these results, a combination of three factors isnecessary to take into account during the design to reject the direct effect of CM
currents generated by power supplies. First, a low level of CM emission at theoutput of the power supply should be specified. Second, a power cable with screen
should be used to limit the radiation of the long cable and finally an appropriate
CM filter has to be included at the input of the FEE. The location of this filter is very
critical, the best position is as close as possible to the entrance of the power cable
into the shielding cabinet or box.
It is important to analyze the cross-effect from CM to DM because the front-end
electronics is more sensitive to CM perturbations coming through the power cables
than DM perturbations. The CM-DM transfer functions are defined as.

I dm ( L )V dm ( L )TFcm – dmI = ---------------- and TF cm – dmV = -----------------I cm ( 0 )V cm ( 0 )(4.87)

Figure 4.28 shows the transfer function for both modes of propagation.

Figure 4.28: CM to DM conversion current and voltage transfer function of HCAL power
supply distribution system - Simulated values.
The current transfer function increases with the frequency. This effect is generated
by the unbalance of the cable impedance at high frequency. On the other hand the159

4. Noise Propagation

voltage TF decreases with the frequency because the input filter of the FEE presents
very low impedance, reducing the noise voltage at the input terminals

4.5.6 Differential mode noise propagation
The terminal configurations of the DM circuit presented are characterized by ageneralized Thevenin equivalent, the same as that found in 4.62 and 4.63:

ZS =Z cm1 ⋅ ( Z cm2 + Z dm )Z cm1 ⋅ Z cm2------------------------------------------------ --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dm
Z cm1 ⋅ Zcm2
Z cm2 ⋅ ( Z cm1 + Z dm )--------------------------------------------- -----------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dmΩ

(4.88)The values of Z cm1 , Z cm2 and Z dm1 correspond to the CM and DM filter capacitor
impedance placed at the output of the power supply. In the present study the CM
capacitor of this filter is equal to 5 nF and the differential capacitor is equal to
1 µF .

VS =

V S1

=

V S2

( Z cm2 + Z dm )Z cm1--------------------------------------------- --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dm
Z cm2
( Z cm1 + Z dm )--------------------------------------------- --------------------------------------------Z cm1 + Z cm2 + Z dm Z cm1 + Z cm2 + Z dm⋅

I cm1 ⋅ Z cm1
I cm2 ⋅ Z cm2

Volts(4.89)

Where the I cm1 = 1 and I cm2 = – 1 .

ZL =Where Z L2 = 0 Ω

Z L1 0Ω , V L = 0 V Load level
00 Z L2(4.90)

Z L1 = 150 Ω C = 1 µ F + 0.2 Ω .

Figure 4.29 shows the DM current and voltage transfer function of the system
between the output of the power supply and the input of the FEE.

I dm ( L )V dm ( L )TF dmI = ---------------- and TFdmV = -----------------I dm ( 0 )V dm ( 0 )(4.91)

Figure 4.30 shows the value of the DM to CM conversion transfer function
generated in the circuit. It represents the ratio between DM (current and voltage) at
the input of the cable and CM (current and voltage) at the end of cable.

1604.5. Noise propagation in cables

I cm ( L )V cm ( L )TFdm – cmI = ---------------- and TF dm – cmV = ----------------I dm ( 0 )Vdm ( 0 )(4.92)The values are very similar to the CM ones because the impedances of the circuit for
both modes are practically the same. These slight differences produce small changes
in magnitude and resonant frequencies of the transfer functions.

Figure 4.29: DM current and voltage transfer function of HCAL power supply distribution
system - Simulated values.

Figure 4.30: DM to CM conversion current and voltage transfer function of HCAL power
supply distribution system - Simulated values.

161

4. Noise Propagation

4.6 Shield models
4.6.1 Surface transfer impedance and admittance
The model presented in section 4.11 does not include the coupling effect in thecentral conductors of currents flowing through the shield. The equivalent circuit of
figure 4.31 represents this coupling mechanism based on the MTL theory for a
sector of line of infinitesimal length [12]. The circuit consists of the inner systemrepresenting the central conductor of the coaxial cable and the braided shield as
reference conductor. The outer system (shield- metallic tray) is considered as atransmission line, where the metallic tray is the reference of the outer system andthe shield is the conductor.

I2

I0 * Zt2

I1

L 20

R2
R1

Inner Conductors

C 12

L12

V 0 * Yt 2

V2
V1

L10

I 0 * Z t1

C 20

C 10

V 0 * Yt 1
Inner face of shield

I0
V0

E xternal face of shield

RS

LS

CS
G round∆l

Figure 4.31: Distributed MTL model for shielded twisted pair cable.
The voltage and the current sources at each inner conductor represent theinteraction between both inner and outer systems. Z t and Y t represent the surface
transfer impedance and the surface transfer admittance. The transfer impedance ofa shield Z Ti is defined as the ratio between the voltage per unit length generated by
the circuit formed by the shield and the conductors inside the shield and the shieldcurrent. The transfer impedance gives the open-circuit voltage developed between
the internal conductors and the shield for one ampere of shield current, in a
cable1-meter long. On the other hand, the transfer admittance of a shield Y Ti isdefined as the ratio between the current per unit length in the conductors inside the1 See figure 4.2
162

4.6. Shield models

shield and the voltage between the shields and the external structure. The transfer
admittance gives the short-circuit current induced in the internal conductors (whenthe internal conductors are shorted to shield) for one volt between the shield andthe external structure, in a cable of 1 meter long. Both surface transfer impedance
and admittance are characteristic parameters of the shielded cables.
The transfer impedance depends on three parameters Z d , M h and M b as definedby equation [13] -[17].

Zt = Zd ( ω ) + j ⋅ ω ⋅ ( M h ± Mb )(4.93)Z d ( ω ) - The diffusion coupling component is due to skin effect in the shield. It is
predominant at low frequencies.

1(1 + j) ⋅ T ⁄ δ
Z d ( ω ) = ---------------------------------- ⋅ ----------------------------------------2 ⋅ π ⋅ a ⋅ σ ⋅ T sinh ( 1 + j ) ⋅ T ⁄ δ(4.94)

where a is the radius of the shield, T is its wall thickness, σ is the conductivity ofthe shield, permeability δ is the skin depth in the shield given by:1
δ = ----------------------------- .
π⋅f⋅σ⋅µ(4.95)

M h - The aperture coupling component is defined as the coupling through theholes of the shield. It plays an important role in the value of the transfer impedanceat high frequencies.
M b -The braid inductance component is defined as the coupling between theexternal and internal layer of the shield.
Figure 4.32 shows the variation in frequency of the surface transfer impedance ofthe HCAL power cable.

Figure 4.32: Transfer impedance of Aluminium and copper shielded HCAL power cable.

1634. Noise Propagation

4.6.2 Solution for the cable including the shield in the frequency
domainThe mathematical model defined by equation 4.1 and 4.2 is complemented with thegenerators VS h ( z, t ) = Z t ⋅ I 0 ( z, t ) and I Sh ( z, t ) = Y t ⋅ U 0 ( z, t ) to include the effect of
the surface transfer impedance and admittance in the inner conductors, as it isshown in figure 4.31 and equations below [4].

∂∂V ( z, t ) = – R ⋅ I ( z, t ) – L ⋅ I ( z, t ) + Zt ⋅ I 0 ( z, t )∂z∂t(4.96)∂∂I ( z, t ) = – G ⋅ V ( z, t ) – C ⋅ V ( z, t ) + Y t ⋅ U 0 ( z, t )∂z∂t(4.97)The solution of the complete equation system starts with the solution of the outersystem, which is approximated by a transmission line model. The external system issolved following the procedure described in section 4.4, and the distributed voltage
Vˆ 0 ( z ) and current ˆI 0 ( z ) at every location z of the outer system (shield - ambient)
are calculated using the equation 4.55 and 4.56.
These voltages and currents are used to calculate the magnitude of the additional
generators defined in equation 4.96 and 4.97. Defining Vˆ Sh ( z ) = ˆI ( z ) ⋅ Z and0tˆI ( z ) = Vˆ 0 ( z ) ⋅ Y , these equations, can be re-written in state-variable form as aShtcoupled set of first order ordinary differential equations given by:ˆˆˆd Vˆ ( z )= 0 – Z ⋅ V ( z ) + V Sh ( z )dt ˆˆI ( z )ˆI Sh ( z )– Yˆ 0I( z)(4.98)The solution to the phasor MTL equation 4.98 can be obtained by the analogy to thestate-variable equations for lumped systems show in equation 4.38. Therefore, thesolution to the phasor MTL equations becomes, by the directly analogy:zˆXˆ ( z ) = Φ̂ ( z – z o ) ⋅ Xˆ ( z o ) + ∫ Φ̂ ( z – τ ) ⋅ V Sh ( τ ) ⋅ dτ(4.99)zo
ˆI Sh ( τ )ˆwhere Xˆ ( z ) = V ( z ) and the chain parameter matrix, Φ̂ ( z ) was previouslyˆI ( z )defined by equation. 4.38. It can be observed that the shield effect adds a

convolution term to the usual chain parameter relation. The solution at z=L ofequation 4.99 is:164

4.6. Shield modelsVˆ ( L ) = Φ̂ 11 ( L ) Φ̂ 12 ( L ) ⋅ Vˆ ( 0 ) + Vˆ ShT ( L )ˆI ( L )ˆI ( 0 )ˆI ShT ( L )Φ̂ 21 ( L ) Φ̂ 22 ( L )(4.100)where the total source voltages Vˆ ShT i ( L ) and currents ˆI ShT i ( L ) are calculated
according to the equation:
LVˆ ShT ( L ) = ∫ [ Φ̂ 11 ( L – τ ) ⋅ Vˆ S h ( τ ) + Φ̂ 12 ( L – τ ) ⋅ ˆIS h ( τ ) ] dτ(4.101)0

LˆI ShT ( L ) = [ Φ̂ 21 ( L – τ ) ⋅ Vˆ ( τ ) + Φ̂ 22 ( L – τ ) ⋅ ˆI ( τ ) ] dτShSh∫

(4.102)0

Based on these equations, the equivalent circuit of the lines - shields system can beviewed as an unexcited line in series with equivalent sources Vˆ ShT i ( L ) and ˆI ShTi ( L ) ,located at z = L as it is shown in figure 4.33.

Figure 4.33: Representation of MTL of a shielded cable having lumped sources that
represents the effects of shield currents and voltages
Now that the general solution of the phasor MTL equations has been obtained interms of the chain parameter matrix and the surface transfer impedance and surface
transfer admittance forcing functions, the terminal conditions to arrive to an explicit
solution for the phasor line voltages and currents are incorporated.
Based on figure 4.33, the voltages and currents at the right end of the unexcited line
are denoted as Vˆ ′ ( L ) and ˆI ′ ( L ) . These are related to the actual desired voltages andcurrents, Vˆ ( L ) and ˆI ( L ) as:Vˆ ′ ( L ) = Vˆ ( L ) – Vˆ ShT ( L )(4.103)

ˆI ′ ( L ) = ˆI ( L ) – ˆI ( L )ShT(4.104)165

4. Noise Propagation

In this way it is possible to incorporate the shield effects in the solution by replacingVˆ ( L ) and ˆI ( L ) in the equations1 for the terminal responses without shield effects
given in the section 4.4.1 with Vˆ ( L ) – Vˆ ShT ( L ) and ˆI ( L ) – ˆI ShT ( L ) . When the generalsolution given in 4.100 in terms of the chain parameter matrix with sources isreformulated in the form of the chain parameter relation without shield effects, it
gives:Vˆ ( L ) – Vˆ ShT ( L ) = Φ̂ 11 ( L ) Φ̂12 ( L ) ⋅ Vˆ ( 0 )ˆI ( L ) – ˆI ShT ( L )ˆI ( 0 )Φ̂ 21 ( L ) Φ̂22 ( L )(4.105)

Considering the terminal conditions written in the form of generalized Thevenin
equivalent:Vˆ ( 0 ) = VˆS + ZˆS ⋅ ˆI ( 0 )(4.106)Vˆ ( L ) = VˆL + Zˆ L ⋅ ˆI ( L )(4.107)

Substituting into equation 4.104 and 4.103 gives:[ Φ̂ 11 ⋅ Zˆ S + Zˆ L ⋅ Φ̂ 22 – Φ̂ 12 – ( Zˆ L ⋅ Φ̂ 21 ⋅ Zˆ S ) ] ⋅ ˆI ( 0 ) =[ Φ̂ 11 – Zˆ L ⋅ Φ̂ 21 ] ⋅ Vˆ – Vˆ L + [ Vˆ ( L ) – ( Zˆ L ⋅ Vˆ ( L ) ) ]S

ShT(4.108)ShTˆI ( L ) = ˆI ( L ) + Φ̂ 21 ⋅ Vˆ S + ( ( Φ̂ 22 – ( Φ̂ 21 ⋅ Zˆ ) ) ⋅ ˆI ( 0 ) )ShTS(4.109)From these values and equation 4.106 and 4.107 the voltages at both ends of thelines can be obtained.
All of these equations are implemented in a MATLAB program to analyze thesusceptibility of the HCAL power cable to shield currents. The validation of themodel presented in this section and a more detailed analysis is shown in appendix
A.

1 Equations 4.66 and 4.67
166

4.6. Shield models

4.6.3 Susceptibility of HCAL power cable to shield currents
4.6.3.1 Introduction
In this section, the influence of the cable’s shield quality in the coupling andpropagation of the noise induced by shield currents in the HCAL power
distribution system is presented. These currents are originated by near and far
electromagnetic fields and by voltage differences at the connection points of thecable’s shield. Figure 4.34 shows a schematic of the circuit under study. Two kindsof shields, aluminum sheet and copper braid have been analyzed. Their surface
transfer impedance has been depicted in figure 4.32.

Vdm , Vcm (0) & I dm , I cm (0)Vdm , Vcm ( L ) & I dm , I cm ( L)Cable
EMI Filter

FEE

VGnd
Figure 4.34: HCAL power supply system layout for shield currents analysis.
The example presented corresponds to a 15 meter long power cable (external radius
6 mm) placed 10 mm above a metallic tray, while a voltage difference is applied
between the two ground connections of the shield at the detector and the power
supply. This cable connects the FEE and the power supply, which are represented in
the presented model by their input and output filters, respectively.
The circuit is split in two parts, inner and outer system. The outer system represents
the external circuit that the cable’s shield forms with the metallic tray. This circuit isapproximated as a transmission line, where the metallic tray is the reference of theouter system and the shield of the conductor. The circuit transmission line
parameters are calculated using the expression of the inductance and capacitancefor a 6 mm external radius cable placed 10 mm above the reference tray, which arepresented in detail in [4]:µhµH
L Sh = ---------- ⋅ acosh ⎛ ---⎞ = 0.22 -------⎝ r⎠2⋅πm(4.110)167

4. Noise Propagation

2⋅π⋅ε
pF
C Sh = ----------------------- = 50.5 ------mhacosh ⎛ ---⎞
⎝ r⎠

(4.111)

The resistance is measured at DC and to include the skin effect in the model theresistance1 (copper braid shield) is represented in the frequency domain as:f
R Sh = ⎛ 0.0035 ⋅ ⎛ 1 + ------------------⎞ ⎞⎝⎝160000⎠ ⎠(4.112)The terminal configurations of the circuit presented are characterized as a
generalized Thevenin equivalent as in 4.62 and 4.63, where:

Z S = 0 Ω , VS = 1 V Power supply level(4.113)Z L = 0 Ω , V L = 0 V Load level(4.114)The second part of the circuit is the inner system representing the central
conductors of shielded cable taking as reference conductor the braided shield. The
per unit length parameter matrices for this cable are based on the measurements,
which has been presented2 in section 4.5.1. The terminal configurations of the circuit
presented are characterized as a generalized Thevenin equivalent, where:

ZL =with Z L2 = 0 Ω

Z L1 0Ω , V L = 0 V Load level
00 Z L2(4.115)

Z L1 = 150 Ω C = 1 µ F .

The equivalent initial configuration of the power supply filter is calculated based on
Kenelly’s Theorem to simplify the calculation of the source impedance matrix. The
final impedance matrix at source level is equal to:

ZS =

1
1
--------------------- + --------------------j ⋅ ω ⋅ C a j ⋅ ω ⋅ Cc

1--------------------j ⋅ ω ⋅ Cc

1--------------------j ⋅ ω ⋅ Cc

1
1--------------------- + --------------------j ⋅ ω ⋅ Cb j ⋅ ω ⋅ Ca

Ω , and V S = 0 V Source level (4.116)
0where C a = 2.005 µ F , C b = 2.005 µ F and C c = 10.025 nF .

mΩ
1 Aluminum sheet shield - DC resistance of 6 --------- and f o ≈ 10 kHzm
2 Equations 4.70 and 4.71
168

4.6. Shield models

4.6.3.2 Common mode
Figure 4.35 shows the trans-admittance between the CM current at the input of theload (z = L) and the CM voltage applied between both ground connections, definedas:

I cm ( L )Y cm = ---------------V Gnd(4.117)

Figure 4.35: CM admittance of shielded (aluminum and copper) HCAL power cable Simulated values.
When a voltage difference is applied between the ground connections, a current
flows through the shield and a CM voltage appears in the central conductors
induced by the transfer impedance. At high frequency, the induced CM voltage andresulting CM currents increase due to the increasing characteristic of the surface
transfer impedance. Nevertheless, at high frequency the shield current decreases
because of the inductive characteristic of the external shield circuit, andadditionally, the resonant frequencies of the cable start to play an important role.
The most important point of this study is the big difference between both kinds ofshields. The aluminum shield has higher admittance values and hence provides
lower noise immunity. As a consequence the final selection of the shield will
depend on the susceptibility of the FEE to this noise currents. In case the FEE is very169

4. Noise Propagation

sensitive to noise, it will be necessary to use a very good shielding to improve thenoise susceptibility of the system.
Figure 4.36 shows the transfer function between the CM voltage at the input of theload and the CM voltage applied between ground connections. This transfer
function is defined as:

V cm ( L )TFcm = ----------------V Gnd(4.118)Figure 4.36: CM voltage transfer function of shielded (aluminum and copper) HCAL
power cable - Simulated values.
As it is expected there is a difference between both types of cables. In addition thevalues of the transfer function for both cases do not decrease with the frequency
because of the real capacitors at the input of the FEE.
4.6.3.3 Differential mode
Figure 4.37 shows the trans-admittance between the DM current at the input of theload and the CM voltage applied between ground connections, defined as:

I dm ( L )Y dm = ---------------VGnd

170(4.119)

4.6. Shield models

Figure 4.37: DM admittance values of shielded (aluminum and copper) HCAL power cable- Simulated values.
Again, there is an important difference between both types of shields. The value ofthe admittance increases with frequency. When a current flows through the shield,
it induces a CM voltage in the central conductors, giving a similar relation to the
CM to DM conversion previously analyzed. This conversion depends on theunbalanced impedance of the system, which at high frequency is caused by thecable inductance imbalances.
Figure 4.38 shows the transfer function between the DM voltage at the input of theload and the CM voltage applied between both grounds.

V dm ( L )TF dm = -----------------V Gnd(4.120)Figure 4.38: DM voltage transfer function of shielded (aluminum and copper) HCAL
power cable - Simulated values.

1714. Noise Propagation

The quality of the shield is very important to define the amount of noise that can becoupled to the central conductors. The value of the transfer function do not decreaseat high frequency due to the real capacitor at the input of the FEE.

4.7 Summary
Based on a multi-conductor transmission line model, the important parameters toanalyze the behavior of both the tracker multi-service cable and the tracker low
impedance cable have been estimated during the first part of this chapter. These
measurements show the dependence of both inductance and resistance values withfrequency. The measurements are used to parameterize the models defined to study
the effects of noise propagation in long cables.
The second part of the chapter addresses the effect of long cables on the noisepropagation between the power supply units and the FEE. CM and DM noise
currents flowing into the input power terminals of the FEE degrades the noise
performance of the sensitive electronics. To avoid that these currents flow into theFEE, CM/DM filters have to be placed at the power entrance. Input filters balance
the load presented to the cable at high frequency, which effectively decreases the
CM to DM and DM to CM conversion. However, these filters amplify the CM
currents flowing through the cable and it can radiate severely. To attenuate thiseffect, power supplies have to exhibit, at the output, a low level of conducted CM
emission and the cables have to present shielding. The quality of the shielded cableand the property of coupling shield currents into the central conductors is definedby the surface transfer impedance. The use of copper braid shields helps to decrease
the coupling of noise due to shield currents.

4.8 References[1] V.K. Tripathi and J.B. Rettig, "A Spice Model for Multiple Coupled Microstrips andOther Transmission Lines", IEEE Transactions on Microwave Theory and Techniques,Vol 33, pp1513 - 1518, December 1985.
[2] A. R. Djordjevic, T. K. Sarkar and R. F. Harrington, "Time domain Response ofMulti-conductor Transmission Lines", Proc. IEEE PROC-75, N-6, pp. 743-764, June
1987.
[3] C.R. Paul, "Introduction to Electromagnetic Compatibility" NY: Wiley-Interscience,
1992, ISBN-0-471-54927-4[4] C.R. Paul, "Analysis of multi-conductor transmission lines", 1992, ISBN-0-471-02080-X

172

4.8. References[5] Lech Znamirowski, Olgierd A. Palusinski, "Algorithm for Accurate CapacitanceMatrix Measurements of the Multi-conductor Structure for VLSI Interconnections",
Private note.
[6] Lech Znamirowski, Olgierd A. Palusinski, "Analysis of error propagation incomputing the Maxwell matrix of multi-conductor structure from measurements", Proc.
IEEE Science, Measurement and Technology, Vol 145, pp 117 - 121, May 1998.
[7] Roger B. Marks, Dylan F. Williams, "Characteristic impedance determination using
propagation constant measurement", IEEE Microwaves and guided wave letters, Vol 1
N-6, pp 141 - 143, June1999.
[8] Sergio A. Pignari, A. Orlandi "Long-Cable effects on conducted emissions levels",IEEE Transactions on Electromagnetic Compatibility, Vol 45, N-1, pp 43-54 February
2003.
[9] S. Canigia, F. Maradei, "Equivalent circuit models for the analysis of coaxial cables
immunity", Proc. IEEE International Symposium on Electromagnetic Compatibility,
pp 881-886, August 2003.
[10] A. Orlandi, "Frequency and time domain modeling of the transfer impedance anddistributed longitudinal induced voltage by means of a Spice equivalent circuit", IEEE
Transactions on Electromagnetic Compatibility, Vol. 45, N-1, pp 125-129, February
2003.
[11] F. Arteche, C. Rivetta, F. Szoncso, "EMC immunity test results of HCAL FEE",
Proc. 9th Workshop on Electronics for LHC experiments, LECC2003, Amsterdam,
pp 358-363, October 2003.
[12] Lorenz-Jung, Jan Luiken ter Haseborg, "Evaluation of measured complex transfer
admittance for the characterization of shield in-homogeneities of Multi-conductor Cables",IEEE Transactions on Electromagnetic Compatibility, Vol. 41, N-4, pp 358-363,
November 1999.
[13] Edward F. Vance, " Coupling to shielded cables ", 1987, ISBN 0-89874-949-2.
[14] Jasper J. Goedbloed, "Electromagnetic Compatibility", 1990, ISBN-0-13-249293-8.
[15] Tyni M., "The transfer impedance of coaxial cables with braided outer conductor",
Proc. Nauk, Inst Telekomun Akust. Politech Wroclaw Ser. Konfi, pp 410-419, 1975.
[16] S.Sali, "An improved Model for the Transfer Impedance Calculations of Braided
Coaxial Cables", IEEE Transactions on Electromagnetic Compatibility, Vol. 33, N-2,
pp 139-143, November1991.
[17] R. De Leo, G. Gerri, V. Mariani Primiani, R. Botticelli, "A simple but effective way
for cable shielding measurement", IEEE Trans. on Electromagnetic Compatibility, Vol.
41, N-3, pp 175-179, August1999.

173

4. Noise Propagation

174

Chapter 5
Susceptibility at early design stage

The purpose of this chapter is to present the assessment of the electromagnetic
compatibility during the early stage of the electronic design. It is partially based on
the models presented in previous chapter. As an example, the connection between
the FEE and the photo-detector of the Forward Hadron Calorimeter is studied. This
area of the HCAL sub-system has the particularity that the read-out amplifiers areplaced 4 meters away from the detector. This characteristic as well as the functional
properties of the QIE, which only admits positive currents, creates one of the mostdelicate connections of the detector, susceptible to very low EMI signal coming fromground currents or external electromagnetic fields. The goal of this chapter is topredict grounding and shielding problems and estimate the effect of the noiseinterference in the electronic system to achieve a design with high noise immunity.

5.1 Introduction
The FEE of Forward Hadron Calorimeter1 is composed of about 3000
photo-multipliers (PMT) arranged in boxes housing 24 PMTs each one. Read-out
amplifiers are arranged in 6 channel daughter cards located about 4 meters from the
PMTs. Photo-detectors are biased using a resistive divider and its gain can be5
6adjusted between 4 ×10 and 5 ×10 . High voltage power supplies are located into
the counting room about 120 m. away from the metallic boxes housing the PMTs.
The anode current is amplified and digitized by a special gated charge amplifier.
This ASIC chip, denominated Charge Integrator Encoder (QIE)2 is a differential
current mode amplifier with an input impedance of either 92 Ω or 50 Ω for thefrequency span of 40MHz and the full input current range. The sensitivity of thisamplifier is 2.7fC/LSB and it integrates the signal during a period of 25nsec. The
dynamic range of each input current is asymmetric, positive currents in both inputs
1 See section 1.7
2 See section 1.7.2 and 1.7.3
175

5. Susceptibility at early design stage

can be processed up to a maximum charge of 27pC, while negative currents canreach only a few fC. Due to the asymmetric dynamic range, two cables with theirown current return have to be used to complete the differential topology. For thepresent study coaxial cables (50 Ω ) have been chosen to connect the PMTs to theamplifiers. The complete circuit for one channel is depicted in figure 5.1

PMT

QIE sig
Va+

Anode
Cp

QIE ref
V a−

Cc
QIE
HV

RGND

Figure 5.1: Circuit Layout of PMTs - QIE
Amplifiers are grouped in 6 channel cards and more than 10 boards are housed insmall crates. All the amplifier cards are locally grounded at the detector ground.
The box holding the PMTs is also locally grounded at the detector structure but
about 4 mts. from the ground connection of amplifiers. As it is depicted in figure5.1, each return point of the PMTs is isolated from the ground connection throughthe resistor RGND .

5.2 Considerations about the noise in the FEE
The noise level defines the minimum signal that the FEE can process. The noise at
the FEE is due to the contribution of different sources and perturbations as thermal
noise of transistors and resistors, noise picked-up by connections between thesensor devices and the FEE, spurious signals, etc. Taking as a common point theanalogue output stage of the QIE to analyse the total noise, it can be written as:

n a ( t ) = n th ( t ) + n cm ( t ) + n emi ( t ) + …

(5.1)

where n th ( t ) is the thermal noise contribution, n cm ( t ) is the noise due to common
mode signals, and n emi ( t ) represent the contribution due to electromagnetic
interference (EMI). These three terms are the most important in this analysis.
The thermal noise is characterized by the power spectral density176

5.3. Thermal noise

2

2

2

2

2

N th ( ω ) = 2 ⋅ [ T e ( ω ) ⋅ e n ( ω ) + T i ( ω ) ⋅ i n ( ω ) ] ⋅ T amp ( ω )
2

2

(5.2)

2

where e n ( ω ) and i n ( ω ) are the spectral density of the equivalent series andparallel thermal noise generators of the QIE and T e ( ω ) and T i ( ω ) are the transfer
function between the voltage or current thermal noise generators and the voltage orcurrent at the input of the QIE.
The additional noise due to common mode signals and EMI can be mathematically
represented as voltage spectral density by:

N cm ( ω ) = T cm ( ω ) ⋅ T amp ( ω ) ⋅ V cm ( ω )(5.3)

N emi ( ω ) = [ T emi H ( ω ) ⋅ H ( ω ) + T emi E ( ω ) ⋅ E ( ω ) ] ⋅ T amp ( ω )(5.4)

where T cm ( ω ) is common mode transference, T amp ( ω ) QIE transference andT emiH ( ω ) , T emiE ( ω ) output voltage to magnetic field and electric field transference,
respectively.
The criteria generally used to set the total noise contribution in a design is to define
the total noise mainly due to the thermal noise, making any external noise
perturbation much lower than the thermal noise. This means:
2

2

(5.5)

2

(5.6)

〈 na 〉 ≅ 〈 n th〉〈 ( n cm + n emi + … ) 〉 → 0

In order to use this criteria for the design it is necessary to know and minimize
2〈 n th〉 and to define the factors T cm ( ω ) , T emiH ( ω ) and T emi E ( ω ) to reduce theircontribution in the total noise.

5.3 Thermal noise
The QIE is the key element of the FEE. It amplifies, shapes and digitizes theincoming signal from the optical devices. The QIE presents two inputs, one for theincoming signal and the other for a reference signal. The input currents can flow
only into one direction in both QIE input. Both input currents are amplified andintegrated by the first stage of the QIE and then subtracted before the analog todigital conversion takes places. The QIE acts as a difference amplifier where theinput current is integrated during 25 ns. If we define i+ and i- , the QIE input
currents, c is the integrator capacitance, and g ( t ) the impulse response of the177

5. Susceptibility at early design stage

amplifying stage before the integration, the analog signal before the digitalconversion can be expressed as:

1 τ+v a ( t ) = v a – v a = --- ⋅ ∫ g ( t ) * ( i+ ( t ) – i- ( t ) ) dt
c 0
+

(5.7)

-

Where v a is the input voltage of the QIE signal, v a is the input voltage of the
QIE reference, τ corresponds to the 25 ns. integration interval and’*’ denotes the
convolution operation. Using the Laplace transform to evaluate the equation 5.7–(s ⋅ τ ) G ( s ) ⋅ ( I ( s ) – I ( s ) )1–e+v a ( s ) = L ( v a ( t ) ) = ------------------------- ⋅ -------------------------------------------------- = T ( s ) ⋅ ( I + ( s ) – I - ( s ) )
s
c

(5.8)

where G ( s ) = L ( g ( t ) ) . the magnitude of T ( s ) can be evaluated in frequency
domain by replacing s = j ⋅ ω = j ⋅ 2 ⋅ π ⋅ f .
The calculate intrinsic thermal noise contribution at the output of the amplifier, eachsignal path of the QIE amplifier can be modeled by the equivalent circuit depicted
in figure 5.2.

en2

G(f )

∫

v+a

Zin

in2

Zo

Figure 5.2: Equivalent circuit to calculate the thermal noise
The cable connecting the photo-tube and the QIE is 4 m. long and the characteristic
impedance is Z o . The cable is matched by the input impedance of the amplifier.
The opposite end of the cable can be considered open in a first approximation. The
amplifier thermal noise is characterized by two independent sources modeling theseries equivalent noise and the parallel equivalent noise. It has been assumed thatboth sources represent white noise characterized by a constant unilateral22
normalized power spectrum e n and in , respectively.
The impedance presented by the line at the input of the amplifier can be calculated
for the case of infinite impedance at the end of cable as:

Z in = – j ⋅ Z o ⋅ cot ( β ⋅ L )

178

(5.9)

5.3. Thermal noiseωv : is the phase velocity of the cable and ω = 2 ⋅ π ⋅ f . The inputv
current at any QIE input in frequency domain is:where β = ----en ( ω )i n ( ω ) ⋅ Z in
i i ( ω ) = ------------------------ + ------------------------- = [ T e ( ω ) ⋅ e n ( ω ) + T i ( ω ) ⋅ i n ( ω ) ]( Z in + Z o ) ( Z in + Z o )(5.10)The normalized power spectrum at each QIE input is:
2

2

2

2

2

i i ( ω ) = T e ω ⋅ en + T i ω ⋅ i n(5.11)and the normalized power spectrum after the integration assuming the noise
contribution at each input statistically independent is:
2

2

2

2

2

n th ( ω ) = 2 ⋅ [ Te ( ω ) ⋅ en ( ω ) + T i ( ω ) ⋅ in ( ω ) ] ⋅ T ( ω )2(5.12)The transfer functions can be calculated by:

Te ( ω )2

Ti ( ω )2

2

2
1[ sin ( β ⋅ L ) ]1= ------------------ = ------------------------------ = -------------------------------------------------------------2
2
2
Z in + Z o( Zo ) ⋅ ( 1 + [ cot ( β ⋅ L ) ] )Zo(5.13)2

Z in
Z in 22= ------------------ = [ cos ( β ⋅ L ) ]- = -------------------------------------------------------------2
2
Z in + Z o( Z o ) ⋅ ( 1 + [ cot ( β ⋅ L ) ] )(5.14)andω⋅τ 2
ω ⋅ τ⎞ 2⎛ ---------sin ⎛ -----------⎞
sin
2
⎝ 2 ⎠
⎝ 2 ⎠
τ
4 ⋅ G( ω)2T ( ω ) = G ( ω ) ⋅ -- ⋅ ------------------------ = ---------------------- ⋅ -------------------------------22c ⎛ ω ⋅ τ⎞cω----------⎝ 2 ⎠In this development Z in + Z o22= Z in + Z o

2(5.15)

because we are assuming that Z o is

β also real. Assuming the bandwidth of the1
amplifying stages is much larger than f = --- , it is possible to assume thatτG ( ω ) = G = cte . Evaluating the integral, we get:real and Z in is imaginary, being

2 2⎛ G ⋅ enτ⎜ ----------------------- ⋅ --⋅ τ⎞ 2
2 2
⎛ sin ⎛ ω⎞224
⎜ c ⋅ Z----------2 ⋅ G ⋅ en ∞ ⎜
⎝ 2 ⎠
2⎟
⎜
o
------------------------------ ⎜ --------------------------------- ⋅ [ sin ( β ⋅ L ) ] ⎟ dω = ⎜
2
2
2
2
2 o ⎜
⎟
⎜G ⋅e
ω
π ⋅ c ⋅ ZonL⎝⎠
⎜ ---------------------- ⋅ ---------⎜ 22 2⋅v⎝ c ⋅ Zo∫ω⋅τ ⎞if ⎛ ( β ⋅ L ) ≥ -----------⎞ ⎟
⎝
2 ⎠ ⎟
⎟
⎟
⎟
ω⋅τ
if ⎛ ( β ⋅ L ) < -----------⎞ ⎟
⎝
2 ⎠⎟
⎠

(5.16)

179

5. Susceptibility at early design stage

2 2⎛ G ⋅ in τ⋅ τ⎞ ⎞⎛(β ⋅ L) ≥ ω
2⎜ -------------------- ⋅ --⎟if---------ω
⋅
τ
⎛
⎝
2 2
⎛
⎞
2
4
2 ⎠ ⎟
⎜
-⎞⎠
2 ⋅ G ⋅ i n ∞ ⎜ sin ⎝ ---------c22⎟⎜⎟---------------------------⎜ --------------------------------- ⋅ [ cos ( β ⋅ L ) ] ⎟ dω = ⎜⎟2 2 τ–L
2
2
-⎜
⎟
o
⎜G ⋅i
⎟
π⋅c
ω
v
ω
⋅
τ
n
⎝
⎠
⎜ ----------------------- ⋅ ------------if ⎛ ( β ⋅ L ) < -----------⎞ ⎟
⎝
2
⎜ 2
2 ⎠⎟
2
⎝ c ⋅ Zo⎠∫

(5.17)

Then the thermal noise is given by the following expression,
⎛
⎜
⎜
⎜
2
〈 n th 〉 = ⎜⎜
⎜ 2
⎜G
-⋅
⎜ -----⎜ c2
⎝

2
⎞
2 ⎛ e
2
τ
G ⎜ n
------- ⋅ ------------ + i n ⎟ ⋅ --⎟ 422 ⎜c ⎝ Zo
⎠

⎞
⎟
⎟
⎟
⎟
⎟
⎟
ω ⋅ τ⎞ ⎟⎛---------if ( β ⋅ L ) <⎝2 ⎠ ⎟⎟⎠

ω⋅τif ⎛ ( β ⋅ L ) ≥ -----------⎞
⎝
2 ⎠

⎛e 2
⎞
2
2 τ
L
n
⎜ ----------- – i n ⎟ ⋅ ------------ + i n ⋅ --⎜ 2
⎟ 2⋅v
2
⎝ Zo
⎠

(5.18)⋅τ
From these equations it is important to note if ( β ⋅ L ) ≥ ω----------- , the length of the cable2
has not any influence on the output power noise, as it is shown in figure 5.3. In
addition, the output power noise decreases if the characteristic impedance of cableincreases.

nth2

ω ⋅τ
2⋅β

L

Figure 5.3: Output power noise - Zo = cte - G = cte
Assuming a propagation velocity of cable equal to 66% of the light speed, thecritical length results equal to 2.5 meters, which results smaller than the cable lengthof the circuit under study.

180

5.4. Susceptibility to CM currents

5.4 Susceptibility to CM currents
The common mode rejection of the differential topology has been studied
considering the circuit depicted in figure 5.4

PMT

QIE sig

Cp

Cc

QIE ref

R gnd
V cm

Figure 5.4: Equivalent circuit to analyse the common mode rejection of the system
The effect of unbalances in the circuit are studied considering a mismatch between
the parasitic capacitance of the PMT anode, socket and board, Cp and thecompensation capacitance Cc. In addition, the relative position of the coaxial cables
respect to the reference tray has been considered, assuming the coaxial cables may
be placed at different heights respect to the reference. For all those cases, the resistor
R GND is changed and used as a parameter to quantify the common mode rejectionof the front-end electronics.
The EMI immunity study of this circuit has been carried out following a similar
analysis to that presented in chapter 4. The solution of this circuit is based on MTL
models [1][2][3] presented in equations 4.96 and 4.97. For that purpose, the circuit issplit in two parts [4][5], inner and outer system as it has been shown in section 4.6.1.
The outer system represents the external circuit that the cables forms with themetallic tray. This circuit is approximated in this example by a transmission line,
where the metallic tray is the reference of the outer system and the shield of the
conductors are the transmission lines.
The per unit length parameters1 of the MTL equations not only can be measured as
it has been shown in section 4.3 but they can also be obtained directly from thedefinition. It can be achieved solving equations 4.3 and 4.7 analytically for simple
topologies or using finite element analysis programs2 for more complex topologies.
The transmission line parameters of the outer part of the presented circuit can be

1 See equation 4.6, 4.8 and 4.14
2 MAXWELL 2D-3D

181

5. Susceptibility at early design stage

simplified to the case of 2 wires (shields) above and parallel to an infinite, perfectly
conducting plane (metallic tray). The per-unit length inductance, capacitance andresistance matrix equations applied in this example can be found in several books[2][3]. A simple diagram of the circuit used to estimated the per unit lengthparameters of the circuit is shown in figure 5.5.

Figure 5.5: Illustration of the calculation of per-unit length inductance of two conductors
above a ground plane.
The expression of the inductance and capacitance of two cables placed h mm above
the ground is given by:µ2⋅h
L ii = ---------- ⋅ ln ⎛ ----------⎞⎝ ri ⎠
2⋅π(5.19)⎛ 4 ⋅ hi ⋅ hj⎞µ
L ij = ---------- ⋅ ln ⎜ -------------------+
1⎟4⋅π⎝ s ij2⎠(5.20)

The capacitance matrix is calculated using the fundamental relationship obtained
for a homogeneous medium., which in this case is considered the free space.:C = µ⋅ε⋅L–1(5.21)In this way it has been calculated the inductance and capacitance matrix of a systemformed by two coaxial cables (external radius (r) 2mm) placed 7 mm (h=9 mm.)above the ground and with the conductor centers separate a distance of 6 mm are:
–6 H
L = 0.4394 0.2302 × 10 ---m0.2302 0.4394(5.22)

F
34.86 – 18.27 × 10 –12 --m– 18.27 34.86(5.23)

C =182

5.4. Susceptibility to CM currents

The inductance and capacitance matrix of a system formed by two coaxial cables
one placed 7 mm above the ground and the other placed 10 mm are:
–6 H
L = 0.4394 0.2360 × 10 ---m
0.2360 0.4969(5.24)

33.95 –16.12 × 10 –12 ---fm– 16.12 30.02(5.25)C =The resistance matrix, which include the shield skin effect1 (without considering theresistance of the metallic tray) is shown as:

R Sh =⎛⎞⎞⎛
f⎜ 0.014 ⋅ ⎜ 1 + ---------------------⎟ ⎟3⎝⎝
250 ×10 ⎠ ⎠

0

0

⎛
⎛
⎞⎞
f
⎜ 0.014 ⋅ ⎜ 1 + --------------------3-⎟ ⎟⎝⎝250 ×10 ⎠ ⎠Ω(5.26)

Once the per unit length parameter are calculated, it is necessary to fix the terminal
configurations to solved the MTL equation in the frequency domain, following theprocedure showed in section 4.6.2, the terminal configurations of the external circuit
presented are characterized as a generalized Thevenin2 equivalent where:

Z L = 0 0 Ω , V L = 0 V Load level
00
0

ZS =

RGND R GND
RGND R GNDΩ , VS =V cm

V Source level(5.27)(5.28)V cm

The second part of the circuit consists of the inner system representing the central
conductor of coaxial cable taking as reference conductor the braided shield. One of
the coaxial cables is used as a signal cable (s) and the other as a reference conductor(r). The per unit length parameter matrix for these cables are based on themeasurements of coaxial cable [6]-[9]. The values of the inductance and capacitance
per unit length matrix are:

L S = L R = 0.25 × 10

–6 H

---m

(5.29)

1 See Appendix I
2 See equations 4.106 and 4.107.

183

5. Susceptibility at early design stage

CS = CR = 100 × 10

– 12 F---m(5.30)In the case of the resistance matrix the resistance of the internal conductors havebeen considered. The shield resistance has not been included in the model as it ismuch lower than the resistance of internal conductor in all frequency range.

R =

⎛
⎞
f
0.0343 ⋅ ⎜ 1 + ---------------------⎟
3⎝
110 ×10 ⎠Ω---m(5.31)The terminal configurations of the internal circuit presented are characterized as a
generalized Thevenin equivalent1:

Z Ls = Z Lr = 50 Ω , V Ls = V Lr = 0 V Load level

1
1
Z Ss = ---------------------- Ω , Z Ss = --------------------Ωj ⋅ ω ⋅ Cp
j ⋅ ω ⋅ Cc

(5.32)

VSs = V Sr = 0 V Source level (5.33)

The simulated values of surface transfer impedance function2 [5][10]-[12] of thecoaxial cable [13]-[15] used in the present analysis is shown below in figure 5.6

Figure 5.6: Simulated values of the surface transfer impedance function.
1 See equations 4.106 and 4.107
2 See equation 4.93
184

5.4. Susceptibility to CM currents
The common mode rejection T cm ( f ) of the complete circuit is defined as:Va
Tcm ( f ) = 20 × log ⎛ ---------⎞⎝ V cm⎠(5.34)

This transference function includes the effects of the cable topology and the QIE
frequency response. The output voltage Va is equal to the difference between the
output voltages of the individual QIE amplifiers, labeled as QIE sig. and QIE ref. in
figure 5.4.
Figure 5.7 depicts the simulated results of the common mode transfer function, for afrequency range between 10kHz and 100 MHz, for different values of RGND
(100 Ω , 1 k Ω , 10 k Ω ), with the parasitic capacitance Cp different of C c . As it canbe observed, the resistor RGND improves the common mode rejection of thesystem proportionally to its magnitude. However, around the cable resonance, theeffect of the resistor is not very important. The common mode transference presents
notches at frequencies 40MHz and 80MHz due to the 25ns. integration interval of
the QIE amplifier.

Figure 5.7: Common mode rejection - Unbalanced circuit in capacitance, Cp=40 pf, Cc=30
pf - Rgnd=100 Ω , 1 k Ω , 10 k Ω both cables at the same height h=7mm.
Let us analyse now the effect of cable positions, unbalances due to the position ofboth coaxial cables respect to the metallic tray reduce the capacity of the system to185

5. Susceptibility at early design stage

reject common mode signals. This effect is due to the currents flowing through theshields are not equal coupling different voltages into the central conductors.
In figure 5.8 depicts T cm ( f ) for two different configurations. In one of them, thesystem has perfectly balanced capacitances Cp and Cc but the coaxial cables areplaced at different heights (7 mm & 10 mm). The other has unbalanced capacitances
but both coaxial cables are placed at the same level. In both cases, the magnitude of
RGND was fixed to 1 k Ω .

Figure 5.8: Common mode rejection - Comparison between the effect of cable position andthe effect of input capacitance unbalanced

5.5 External electromagnetic fields
5.5.1 Introduction
The effect of coupling external electromagnetic fields into a multi-conductortransmission line is also well described in [16]-[18]. The coupling effect inmulti-conductor transmission lines was first introduced by Clyton [19] and later
extended to multi-conductor lines by Harrison [20]. A short description of the mostimportant aspects is presented here.

186

5.5. External electromagnetic fields

The effect of the incident field is configured as distributed sources along the line.
Incident field effects appear as a set of voltage and current sources. Figure 5.9 showsa generic circuit of an infinitesimally length ∆ z illuminated by an external
electromagnetic field, where the parameters per-unit length have been identified.

Figure 5.9: Distributed MTL model for external field
The MTL equations including the external electromagnetic field effects are:∂∂V ( z, t ) = – R ⋅ I ( z, t ) – L ⋅ I ( z, t ) + V F ( z, t )∂z∂t(5.35)∂∂I ( z, t ) = – G ⋅ V ( z, t ) – C ⋅ V ( z, t ) + I F ( z, t )∂z∂t(5.36)

where the Nx1 vectors on the right hand side, VF ( z, t ) and I F ( z, t ) , model theeffects of the incident fields and are determined by the components of the incident
electric and magnetic fields that lie in the transverse plane. They are as wellinfluenced by the geometry of cables respect to the reference. These vectors are
given by:

…
a'

∂
i
Vˆ F ( z, t ) =∂ t ∫ B ⋅ a n dl

(5.37)

a…

187

5. Susceptibility at early design stage…
a'

ˆI F ( z, t ) = – G ⋅…
a'∫ E t ( dl ) – C ⋅ ∫ E t ( dl )iai

(5.38)a…

…

It can be observed, that equation 5.35 and 5.36 are identical to the MTL equations
including shield effects (equations 4.96 and 4.97). Therefore, the solution to thephasor MTL equation, can be obtained by the analogy to the state-variable
equations for lumped systems, as it has been well detailed in the previous sections1.
The system can be divided again into two parts. One unexcited and the effect of theincident field can be represented by several current and voltage sources located at z
= L as it is shown in figure 5.10.

Figure 5.10: Representation of MTL model with incident-field illumination
The incident field effect add a convolution term to the usual chain parameter
relation2. Then, writing equations 4.40 out for a MTL system illuminated by an
external field of total length L gives:Vˆ ( L ) = Φ̂ 11 ( L ) Φ̂12 ( L ) ⋅ Vˆ ( 0 ) + Vˆ FT ( L )ˆI ( L )ˆI ( 0 )ˆI FT ( L )Φ̂ 21 ( L ) Φ̂22 ( L )(5.39)

where the total source voltages Vˆ FT i ( L ) and currents ˆI FTi ( L ) are according to theequation:
1 Section 4.4
2 Equation 4.40
188

5.5. External electromagnetic fields

L

Vˆ FT ( L ) = ∫ [ Φ̂ 11 ( L – τ ) ⋅ Vˆ F ( τ ) + Φ̂ 12 ( L – τ ) ⋅ ˆI F ( τ ) ] dτ

(5.40)0L

ˆI FT ( L ) = [ Φ̂ 21 ( L – τ ) ⋅ Vˆ F ( τ ) + Φ̂22 ( L – τ ) ⋅ ˆI F ( τ ) ] dτ∫

(5.41)

0

Now that the general solution of the phasor MTL equations has been obtained interms of the chain parameter matrix and the incident field forcing functions, theterminal conditions to arrive at an explicit solution or the phasor line voltages andcurrents are incorporated to yield:[ Φ̂ 11 ⋅ Zˆ S + Zˆ L ⋅ Φ̂ 22 – Φ̂ 12 – ( Zˆ L ⋅ Φ̂ 21 ⋅ Zˆ S ) ] ⋅ ˆI ( 0 ) =[ Φ̂ 11 – Zˆ L ⋅ Φ̂ 21 ] ⋅ Vˆ – Vˆ L + [ Vˆ ( L ) – ( Zˆ L ⋅ Vˆ ( L ) ) ]S

FT(5.42)FTˆI ( L ) = ˆI ( L ) + Φ̂ 21 ⋅ Vˆ S + ( ( Φ̂ 22 – ( Φ̂ 21 ⋅ Zˆ ) ) ⋅ ˆI ( 0 ) )FTS(5.43)

From these values and equations 4.62 and 4.63 the voltages at both ends of the lines
can be calculated.

5.5.2 Uniform plane-wave excitation of the line
For the case of radiation of the line by an uniform plane wave from some distantsource, the fields in the far field region of a radiating structure are spherical waves
which locally resemble uniform plane waves. This assumption simplifies theevaluation of the sources in the above results.
In order to characterize the frequency-domain response of the line let us describe
the incident uniform plane wave angle of incidence and polarization with respect toa spherical coordinate system as it is shown in figure 5.11

Figure 5.11: Characterization of incident field as uniform plane wave - Rectangularcoordinate system.

189

5. Susceptibility at early design stage

The polarization of electric field vector is described in terms of the relation to theunit vectors in the spherical coordinate system, a θ and a φ . In terms of thesecomponents, the general expression for the phasor electric field vector can be
written as:–( j ⋅ βx ⋅ x )–( j ⋅ βy ⋅ y )–( j ⋅ βz ⋅ z )E i = Eˆ o ⋅ ( e x ⋅ a x + e y ⋅ a y + e z ⋅ a z ) ⋅ e⋅e⋅e(5.44)

where the components of the incident electric field vector along the x,y and z axes of
the rectangular coordinate system describing the line are:e x = sin θ E ⋅ sin θ p(5.45)

e y = ( – sin θ E ⋅ cos θ p ⋅ cos φ p ) – ( cos θ E ⋅ sin φp )(5.46)

e z = ( – sin θ E ⋅ cos θ p ⋅ sin φ p ) + ( cos θE ⋅ cos φp )(5.47)

22

2( e x + ey + e z ) = 1(5.48)

The components of the phase constant, β along those coordinate axes are:β x = – β ⋅ cos θ p(5.49)

β y = – β ⋅ sin θp ⋅ cos φ p(5.50)

β z = – β ⋅ sin θ p ⋅ sin φ p(5.51)The phase constant is related to the frequency and properties of the medium as:1
β = ω ⋅ µ ⋅ ε = ----- ⋅ µ r ⋅ ε rvo(5.52)

where v o is the phase velocity in free space and the medium is characterized bypermeability, µ and permittivity, ε , Êo is the complex amplitude of thesinusoidal wave.
The results of the integral of equations 5.37, 5.38, 5.40 and 5.41 are very complex andthey are well detailed in [3]. However, if the integrals 5.40 and 5.41 are solved as asum of integrals (equation 5.53 and 5.54) in which each part of the line is electricallysmall, several assumptions and simplifications can be done to solve these equations.

190

5.5. External electromagnetic fieldsnVˆ FT ( L ) =li + 1

⎛
l
–l ⎞
i+1 i
Φ̂ 11 ( L – τ ) dτ ⋅ Vˆ F ⎛ ----------------------⎞ ⎟ + ⎜
⎝
⎠
⎟
⎜
2
i
⎠
⎝

⎛
⎜
⎜
⎝

li + 1l–l ⎞ ⎛i+1 i
ˆΦ 21 ( L – τ ) dτ ⋅ Vˆ F ⎛ ----------------------⎞ ⎟ + ⎜
⎠⎟ ⎜
2
i⎝
⎠ ⎝

∑ ∫l ii=1nˆI ( L ) =
FT

⎛
⎜
⎜
⎝

∑ ∫l i

i=1

l
–l ⎞
i+1 i
Φ̂ 12 ( L – τ ) dτ ⋅ ˆI F ⎛ ----------------------⎞ ⎟
⎝
⎠⎟
2
i
⎠

(5.53)

l
–l ⎞
i+1 i
ˆ
Φ 22 ( L – τ ) dτ ⋅ ˆI F ⎛ ----------------------⎞ ⎟
⎠⎟
2
i⎝
⎠

(5.54)

li + 1∫l i

li + 1∫l i

For each portion of the integral, the line is electrically short, L « λ . This
simplification allows to evaluate the values of Vˆ Fi and ˆI F i at any convenient pointon each part of the line resulting this value constant. We have arbitrarily chosen toevaluate them at the midpoint of each portion. In case of one conductor of length L(electrically small at the frequency of interest) which is at a certain distance of its
reference conductor and one conductor above the ground plane, the values ofVˆ F and ˆI F are presented below:
Two conductors:ddLsin ⎛ β x ⋅ ---⎞– ⎛ j ⋅ β x ⋅ ---⎞– ⎛ j ⋅ β z ⋅ ---⎞
⎝
⎝
⎝
2⎠
2⎠
2⎠
ˆV ⎛ --L-⎞ ⋅ L = A ⋅ Eˆ o ⋅ ------------------------- ⋅e⋅e⋅ ( j ⋅ β z ⋅ e x – ( j ⋅ β x ⋅ e z ) ) (5.55)
F⎝ ⎠2⎛β ⋅ d⎞--⎝ x 2⎠ddLsin ⎛ β x ⋅ ---⎞– ⎛ j ⋅ β x ⋅ ---⎞– ⎛ j ⋅ β z ⋅ ---⎞
⎝
⎠
⎝
⎠
⎝
2
2
2⎠
L
ˆI ⎛ ---⎞ ⋅ L = – ( j ⋅ ω ⋅ c ⋅ A ) ⋅ Eˆ o ⋅ -------------------------- ⋅ e⋅e⋅ ( ex )F⎝ ⎠2⎛β ⋅ d⎞--⎝ x 2⎠

(5.56)

where A = d ⋅ L is the area between conductors, being d the distance between
them.
For the case of a single conductor above the ground these sources are:⎛L⎞– j ⋅ β z ⋅ --sin ( β x ⋅ h )⎝2⎠ˆV ⎛ --L-⎞ ⋅ L = 2 ⋅ A ⋅ Eˆ o ⋅ ------------------------⋅e⋅ ( j ⋅ β z ⋅ ex – ( j ⋅ βx ⋅ ez ) )F⎝ ⎠2( βx ⋅ h )⎛(5.57)

L⎞– j ⋅ β z ⋅ --sin ( β x ⋅ h )⎝2⎠
ˆI ⎛ --L-⎞ ⋅ L = – ( j ⋅ ω ⋅ c ⋅ 2 ⋅ A ) ⋅ Eˆ o ⋅ ------------------------⋅e⋅ ( ex )F⎝ ⎠2( βx ⋅ h )(5.58)

where A is the area between conductors and the ground plane (h.L) and h the
distance from the conductor to the reference plane.

191

5. Susceptibility at early design stage

All of these equations and equations 5.42, 5.43, 4.62 and 4.63 have been
implemented in a MATLAB program and they have been used in next section to
calculate the effect of external electromagnetic fields in the PMTs and QIE
connections of the Forward Hadron Calorimeter. The validation of the program andthe model presented is explained in appendix A

5.5.3 Susceptibility to Electromagnetic fields of HF FEE.
As it has been showed above, to include the effect of external electromagnetic fields
in the model of the multi-conductor cable1, the outer system is modified. A series
voltage generator Vˆ F ( z ) proportional to the magnitude of the external magnetic
field and a current generator ˆI F ( z ) proportional to the external electric field areincluded in the outer circuit. The modified circuit is solved in a similar manner thansections 4.6.3, first the shield current is calculated to define the coupling generators
to the inner system and afterwards, the inner circuit is solved.
The example presented in this study assumes that the EM wave is propagating in
the positive direction of z with the electric field polarized in the positive direction ofx and the magnetic field perpendicular to the plane created by the cable and themetallic tray, which has been chosen as a reference plane. The case under study isreferred as electromagnetic plane wave with an end-fire excitation as it is showed infigure 5.12.
x
r
Eˆ i

z

r
Hˆ i

y

Figure 5.12: Electromagnetic plane wave with an end-fire excitation
For the end-fire excitation field, the normalized projections of the electric field
vector on the rectangular coordinate system ( e x , e y , e z ) are equal to e x = 1 andey = ez = 0 .
The distributed voltages and currents in frequency domain V F ( z ) and I F ( z ) arecalculated assuming a plane wave and a constant electric field Eo for all
frequencies. Substituting these considerations into equations 5.57 and 5.58 results⎛z⎞– j ⋅ β z ⋅ --sin ( β x ⋅ h )⎝2⎠zVˆ F ⎛ ---⎞ = 2 ⋅ h ⋅ Eˆ o ⋅ ------------------------- ⋅ e⋅ ( j ⋅ βz )⎝ 2⎠
( βx ⋅ h )1 See section 4.6.1
192(5.59)

5.5. External electromagnetic fields⎛z⎞– j ⋅ β z ⋅ --⎝sin ( β x ⋅ h )2⎠
ˆI ⎛ --z-⎞ = – ( j ⋅ ω ⋅ c ⋅ 2 ⋅ h ) ⋅ Eˆ o ⋅ ------------------------⋅eF⎝ ⎠2( βx ⋅ h )(5.60)The magnitude of the electric field applied is 100 µV/m that corresponds to thelimit of the emission level suggested by the EN 55022-B standard, and h is the cableheight respect to the reference plane.
The per unit length parameter matrices1 and terminal configurations2 for the innerand outer systems are the same as section 5.4, except the terminal configurations of
the external circuit, whose matrices are:

ZL = 0 0 Ω
0 0

ZS =

RGND R GND, VL =

0 V Load level0Ω , V S = 0 V Source level0
RGND R GND(5.61)(5.62)

Figure 5.13 depicts the voltage at the output of the QIE when an electromagnetic
field of 100 µV/m is coupled to the cable connection for different values of theground resistor, R GND , when the capacitances Cp and Cc are not equal.

Figure 5.13: Output voltage of QIE in an unbalanced configuration, Cp = 40pf, Cc=30pf,
for different values of ground resistance R GND =100 Ω , 1 k Ω , 10 k Ω
1 See equations 4.6, 4.8 and 4.14
2 See equations 4.106 and 4.107

193

5. Susceptibility at early design stage

The resistor R GND has minimum influence rejecting this perturbation. This isdifferent to the case of CM currents, where the CM rejection is quite susceptible to
the resistance value. This difference can be explained by the different nature of thecircuits. In the first case the ground resistance is in series with the noise source,
however in the second one is in parallel decreasing a lot the beneficial effect of thisresistance.
The influence of unbalances generated by the different position of the cables on thecable tray degrades the performance of the differential amplifier more than thecapacitance unbalance. The effect of the cable heights is compared in figure 5.14
with the effect of different capacitances for a resistor RGND = 1 k Ω .

Figure 5.14: Output voltage of QIE for two different unbalanced configuration. Different
high and unbalanced capacitances with a constant ground resistance of RGND =1000 Ω

5.6 Summary
This chapter has presented a preliminary noise immunity analysis of the differential
amplifier to be used by the FEE of CMS very forward calorimeter. The study of thesensitivity of the FEE to common mode spurious signals and EMI has been
conducted by simulation using a line model based on the MTL theory. Results haveshown the limitations of the topology to common mode rejection and EMI if it is
implemented with separated coaxial cables. It was shown that the position of thecable respect to the reference plane has strong influence in the rejection of thedifferential amplifier. The rejection to spurious signals can be improved, in thisparticular case, by selecting a double coaxial cable with a single braided shield. This
cable balances the outer circuit impedance for both internal twisted pairs.
1945.7. References

Respect to the effect of the ground resistor RGND , the results show that improves
the common mode rejection of the system proportionally to its magnitude,
decreasing at high frequency. However, this effect is practically neglected with thenoise generated by external electromagnetic fields. This difference between bothcases can be explained by the different nature of the circuits.

5.7 References[1] A.R. Djordjevic, T.K. Sarkar, R.F. Harrington "Time domain response ofMulti-conductor Transmission Lines", Proc. IEEE PROC-75, N-6, pp 743-764, June
1987.
[2] C.R.Paul, "Introduction to Electromagnetic Compatibility", NY: Wiley-Interscience,
1992, ISBN-0-471-54927-4[3] C.R. Paul, "Analysis of multi-conductor transmission lines", 1992, ISBN-0-471- 02080-X[4] Lorenz-Jung, Jan Luiken ter Haseborg, "Evaluation of measured complex transfer
admittance for the characterization of shield in-homogeneities of Multi-conductor Cables",IEEE Transactions on Electromagnetic Compatibility, Vol 41, N-4, pp 358-363,
November 1999.
[5] Edward F. Vance, " Coupling to shielded cables ", 1987, ISBN 0-89874-949-2.
[6] T. Williams, "EMC for Product Designers", 1997, ISBN-0-7506-2466-3[7] C. Goldstein, P. Mani, "CW and pulsed mode transfer impedance measurements incoaxial cables", IEEE Transactions on Electromagnetic Compatibility, Vol. 34, pp 50-57
February 1992.
[8] Thomas Kley, "Electromagnetic coupling to and from shielded cables-optimizing
factors", IEEE Transactions on Electromagnetic Compatibility, Vol. 34, pp 39-46
February 1992.
[9] Thomas Kley, "Measuring the coupling parameters of shielded cables", IEEE
Transactions on Electromagnetic Compatibility, Vol. 35, pp 10-20, February 1993.
[10] Jasper J. Goedbloed, "Electromagnetic Compatibility", 1990, ISBN-0-13-249293-8.
[11] M. Tyni , "The transfer impedance of coaxial cables with braided outer conductor", Pr.
Nauk, Inst Telekomun Akust. Politech Wroclaw Ser. Konfi, pp 410-419, 1975.
[12] S.Sali, "An improved Model for the Transfer Impedance Calculations of Braided
Coaxial Cables", IEEE Transactions on Electromagnetic Compatibility, Vol. 33, N-2,
pp 139-143, November1991.

195

5. Susceptibility at early design stage[13] A.Fourie, O. Givati, A.R.Clark, "Simple technique for the measurement of thetransfer impedance of variable length coaxial interconnecting leads", IEEE Transactions on
Electromagnetic Compatibility, Vol. 40, pp 163-166, February 1998.
[14] A. Morriello, T.M. Benson, A.P. Duffy, C.F. Cheng, "Surface transfer impedance
measurement: a comparison between current probe and pull on braid methods for coaxial
cables", IEEE Transactions on Electromagnetic Compatibility, Vol. 40, pp 69-76
February 1998.
[15] R. De Leo, G. Cerri, V. Mariani Primiani, R. Botticelli, "A simple but effective
way for cable shielding measurements", IEEE Transactions on Electromagnetic
Compatibility, Vol. 41, pp 175-179, August 1999.
[16] Ashok K. Agrawal, Harold J. Price, Shyam H. Gur-baxami, " Transient Responseof Multi-conductor Transmission Lines Excited by Non-uniform Electro-magnetic Fields ",IEEE Transactions on Electromagnetic Compatibility, Vol. 22, N- 2,1980.
[17] S. Tkatchenko, F. Rachidi, M. Ianoz, "Electromagnetic field coupling to a line offinite length: theory and fast iterative solutions in frequency and time domains", IEEE
Transactions on Electromagnetic Compatibility, Vol. 37, N-4, pp 509-518, August
1993.
[18] F. Rachidi, "Formulation of the field-to-transmission line coupling equations in termsof magnetic excitation field", IEEE Transactions on Electromagnetic Compatibility, Vol.
35, N-3, pp 404-407, August 1993.
[19] C.R. Paul., "A SPICE model for multiconductor transmission lines excited by an
incident electromagnetic field", IEEE Transactions on Electromagnetic Compatibility,
Vol. 36, pp 342-354, November 1994.
[20] C.W. Harrison, "Generalised theory of impedance loaded multi-conductortransmission lines in an incident field", IEEE Transactions on Electromagnetic
Compatibility, Vol. 14, pp 56-63, May 1997.

196

Chapter 6
RF Susceptibility Test of FEE

The electromagnetic (EM) immunity [1][2] of a device, equipment or system is its
ability to perform without degradation in presence of EM disturbances. The
characterization of the front-end electronics to these disturbances [3] before thedetector is integrated is highly important to guarantee the design goals and thegood performance of the system. EMC immunity tests allow to determine the levelof sensitivity of the FEE and give the possibility of taking corrective actions at an
early design stage to ensure the good compatibility among the sub-systems during
the integration of the detector. The procedure to conduct the radio-frequency (RF)immunity tests consists in the injection of perturbing currents through the
input/output cables of the FEE to simulate the effect induced by conductedinterference and external EM fields. This chapter addresses the results of thedescribed tests when applied to the FEE of the CMS Hadron calorimeter (HCAL).

6.1 Introduction
The RF immunity tests are based on European standards [4] and aerospace rules.
The goal of these tests is to define the immunity level of the FEE to the conducted
disturbances coupled into the input cables, identify the key elements for the noise
degradation of the FEE and settle on the noise level required at the output of thepower supplies. Noise can be coupled to the FEE through the input power cables
and slow control cables, as these are the two paths for conductive coupling due tothe FEE output signal is transmitted to the counting room via optical fibre. RF noise
currents can be induced in the cable shield by near and far external EM fields,
forced to flow through the central conductors by the power supply outputs, etc. To
emulate these interferences in a test environment, four tests have been proposedand conducted to study FEE immunity level due to inject currents via the input
power cables of the FEE. In each of them the transfer function between the external
perturbing signal and the output signal of the FEE is calculated. These tests are:• Injection of shield currents.

197

6. RF Susceptibility Test of FEE• Injection of CM currents in the central conductors.
• Injection of DM currents in the central conductors.
• Injection of CM+DM currents in the central conductors

6.1.1 Injection of shield currents
High frequency currents flowing through the power cable shield can be generated
by far and near electromagnetic fields coupled to the shield, or by high frequency
ground currents flowing in the system. To emulate its effect on the FEE, a sine-wave
current is injected into the power cable shield. This shield current also couples CM
currents to the internal conductors through the surface transfer impedance andadmitance of the cable. All these currents affect the performance of the FEE and its
effect depends on the amount of noise current that can penetrate into the FEE( I pert ), as it is shown in figure 6.1. This plot shows the distribution of the noise
currents in one of the HCAL RBX. This circuit is used to analyze the effects of thesecurrents in the performance of the FEE. Shield currents are present in the CMS
detector and it is necessary to evaluate the immunity of the system to them.

I Shield
I CM Filter I pert FEE

PS

Vout
I strap'

Cable
Safety Gnd

I Shield
I box
I gnd

I stray1
Rf . Plane

I stray2

Figure 6.1: Shield current distribution into the FEE.
The immunity of the FEE to shield currents is quantified defining the transfer
function between the FEE output voltage ( Vout ) and the current that is able topenetrate into the system, ( I CM + I' shield ) (marked in red)V out
TF = ------------------( I shield )

198

(6.1)

6.1. Introduction

This transfer function will basically depend on the ground connections of thesystem. The connection between the filter box and reference plane is the mostimportant connection to obtain good immunity of the FEE against shield currents as
this is the only path that can be used to filter the noise currents before they reach the
FEE. The system immunity to the RF shield noise current increase when the current
I strap can be by-passed, lowering the effect of the perturbing current I pert in theFEE.

6.1.2 Injection of CM currents in the central conductors
Power supplies and FEE can generate CM currents that flow through the central
conductors of a power cable. These currents can affect the performance of the FEE,
depending on the amount of noise that can penetrate into the FEE ( I pert ), as
mentioned before. In the CMS detector it is intended to install SMPS that have highCM current emission levels, and hence the study of the immunity of the system tothese currents is one of the most important studies to be performed on the FEE.
.

I CM

PS

I CM

FilterI pert

FEE
Vout

Cable
Safety Gnd

I strap

I Shield
I box
I gnd

I stray1
Rf . Plane

I stray 2

Figure 6.2: CM current distribution into the FEE.
Figure 6.2 shows the block diagram of the selected circuit to study the immunity of
the HCAL FEE to CM currents as well as the distribution of the CM currents in the
HCAL RBX. The immunity of the FEE to CM currents is quantified defining thetransfer function between the FEE output voltage ( V out ) and the CM current that isable to penetrate into the system, ( 2 ⋅ I CM ) (marked in red)V out
TF = ---------------2 ⋅ I CM

(6.2)

199

6. RF Susceptibility Test of FEE

This transfer function depends mainly on the ability of the system to filter thesenoise currents at the entrance of the power cables to the FEE box. For CM currents,
the cable shield and the straps are the two parts available to bypass these currents
and hence avoid that this perturbation flows though sensitive areas of the FEE.
Most of the CM currents return through the shield, but the input filter must to
provide a low CM impedance to set the CM circuit around the filter.

6.1.3 Injection of DM currents in the central conductors
Both power supplies and the FEE can generate DM currents that flow through thecentral conductors of a power cable. In the CMS detector, any PS units generate DM
noise and therefore the study of the immunity of the system to these currents is also
very important.
DM currents can affect the performance of the FEE, depending on the amount ofnoise that can penetrate into the FEE ( I pert ), as shown in figure 6.3. This plot shows
the selected circuit to study the distribution of the DM currents in the HCAL RBX
and defines the DM transfer function.

I DM

PS

I DM

FilterI pertFEE
Vout

I strap

Cable
Safety Gnd

I box

I stray 1
Rf . Plane

I stray 2

I gnd

Figure 6.3: DM current distribution into the FEE.
If this plot is analyzed in detail there are two differences when compared with theprevious ones. To increase the CM impedance of the system layout, the groundconnection of the power supply and the normalized common impedance box is
disconnected. In this case, the injected current in the active power cable is forced toreturn through the complementary power cable and the CM noise injected to the
FEE is negligible. This configuration is extremely sensitive to resonances in thecircuit and introduces several limitations when evaluating the DM transfer
function.

200

6.1. Introduction

The immunity of the FEE to DM currents is quantified defining the transfer function
between the FEE output voltage ( V out ) and the DM current flowing through thecentral conductor of the cable ( I DM ) (marked in red).

V out
TF = ---------I DM(6.3)

This transfer function depends mainly to the ability of the RBX to filter this noisy
current as well as the capacitors placed on the backplane at FEE the level, which
practically filter all DM currents that escape the input filter. For this reason theefficiency of the input filter is not very important as this is not the only device that isused to filter the DM noise currents prior to the FEE.

6.1.4 Injection of CM+DM currents in the central conductors
This test is performed to analyze the effect of currents, combining common modeand differential mode components, on the FEE immunity and also to define
normalized specifications for the output noise of the power supplies. Figure 6.4
shows the circuit and the current distribution in the system during this test.

I1 = I CM + I DM

PS

I 2 = I DM − I CM

I1

FilterI pertFEE
Vout

I strap

Cable
Safety Gnd

I Shield

I box

I stray1
Rf . Plane

I stray 2

I gnd

Figure 6.4: Positive lead current distribution into the FEE.
In this example, the immunity of the FEE to the positive lead noise currents isquantified defining the transfer function between the FEE output voltage ( Vout )and the noise current ( I 1 ) in figure 6.4 (marked in red).

Vout
TF = ---------I1

(6.4)

201

6. RF Susceptibility Test of FEE

Assuming the total current flowing in each power cable can be expressed as:

I 1 + I 2 ≅ I shield + I gnd = I T(6.5)

where I 1 and I 2 are the currents flowing through the central conductors, and I shieldand I gnd are the return currents through the shield and the ground plane. The
currents flowing through the central conductors can be decomposed in two
orthogonal components:

I1 + I2
I1 – I 2
I cm = --------------- and I dm = -------------2
2

(6.6)

If the injected current I 1 can be decomposed in two components I cm and I dm ,multiplying each current by the respective TF, and adding the resulting output
voltages, the calculated TF following this method should be equal to the TF
obtained by the present measurement.

6.2 Experimental set-up
An experimental set up was implemented to study the immunity of the FEE. Figure
6.5 shows a schematic of the experimental set-up to define the FEE immunity.

Spectrum
Analyser (Ip)
RF Generator
Noise source

Acquisition
SystemRMS - ENC

CommonImpedanceDig
FEE (12 -channels)
VictimAnalg.

Measurement
current probe

Injection
current probe
Figure 6.5: FEE immunity test set-up.

202

Power Supply

6.2. Experimental set-up

The basic idea of the test set-up is to keep the topology as close as possible to thefinal one. The FEE and the auxiliary equipment are placed on a copper plane, 1
meter above the floor. This copper sheet (2x2 meters) is the reference ground plane.
A Bulk injection current probe (Fischer F-120-9A)1, a RF amplifier (Dressler
250/75A) and a RF signal generator (HP 8664B) are used to inject the perturbing
signal to the FEE, while an inductive current clamp (EMCO-91550-1) and aspectrum analyzer (HP-8594E) are used to monitor the current injected. The output
signal of the FEE is measured by its own acquisition system. The connection
between the bulk injection probe and the amplifier is made though a RF atenuator
to improve the ratio between the injected current and the harmonics generated by
the amplifier in 10dB.
To represent the effect of very long cables, normalized common impedances (CI)(Common Mode and Differential Mode impedance) based on lumped components
are used to standardize the measurement. Figure 6.6 shows the common impedance
used in these test.

Figure 6.6: Common impedance - HCAL power cable.
The common impedance presents a stable and well-defined impedance ( ± 25% )throughout the desired frequency range. The chosen values of the CI components
are estimated following the measurement of the HCAL power cable parameters2.
This CI presents three frequency areas in which the impedance behavior of the CI isdifferent. At high frequency the CM and DM impedance of the CI approaches the
CM and DM characteristic impedance of the lines. The chosen values of thecommon impedance in each branch of the line that match the DM and CM
characteristic impedance of the cable are 21.4 Ω for DM and 30.4 Ω for CM. A
more detailed analysis of the cable characteristic impedance matrix is presented inappendix I. At low frequency the value of this impedance is equal to the resistanceof cable, whereas the transition frequency range between the low and highfrequency is defined by inductance of cable.
1 See appendix C
2 See chapter 4

203

6. RF Susceptibility Test of FEE

The values of each branch of the CI for the HCAL power cable in the range 10 kHz
to 50 MHz are measured with a network analyzer (HP 4195A) and are shown infigure 6.7.

Figure 6.7: Variation of the Nominal and measured values of the CI respect to the frequency- CM and DM configuration.

6.2.1 HCAL FEE prototype
The RF immunity test is carried out on a prototype of the HCAL front-end
electronics. Although it has been described at the beginning of the presented
document1 and in [5], some additional details are presented.
In this prototype, the sensitive amplifier is a charge integrator sampled at 34 MHz(the final version operates at 40 MHz). It amplifies and digitizes the signal
generated by a hybrid photo-multiplier located a few centimeters from theamplifier. Later, the sampled signals are collected, serialized and sent out of thedetector via optical links to the acquisition system. At that location, the FEE output
data are processed and converted into deposited energy value that is stored until
the first trigger level signal arrives.
CM and DM signals are injected through the power cables and the output signals of
12 channels are acquired. These channels are distributed in two identical boards,
each of them connected to the backplane. Figure 6.8 shows the board layout of the
1 Section 1.7
2046.2. Experimental set-up

HCAL-FEE. Analogue signal conditioners and digitization electronics are locatedon these boards, which constitute one of the most sensitive components of the
HCAL electronics.

Ch 1 & Ch 7
Ch 2 & Ch 8
Ch 3 & Ch 9
Ch 6 & Ch 12
Ch 4 & Ch 10
Ch 5 & Ch 11
Board connector

Figure 6.8: HCAL-FEE board layout [6].
The root mean square (RMS) of the acquired data is used to evaluate theperformance of the FEE operating under the perturbing signal. Figure 6.9-a depicts
the output signal in the time domain when no perturbing signal is applied, while
figure 6.9-b shows the histogram of the signal. Figure 6.10 depicts the output signal
and its histogram when the FEE is perturbed by a sine wave current of 10 MHz.

ab

Figure 6.9: Time domain and histogram of the output of channel 2 HCAL - FEE when is
not perturbed.

205

6. RF Susceptibility Test of FEEab

Figure 6.10: Time domain and histogram of the output of channel 11 HCAL - FEE when isperturbed
During the test, the RMS value of the output signal is used to evaluate the noise of
the FEE. A level of 2.16 counts is the target value for this design for the RMS output
voltage corresponding to the amplifier thermal noise. It is equivalent to an
Equivalent Noise Charge (ENC) at the input of the amplifier equal to 0.72 fC or 4500
electrons1. In these prototypes, when no RF perturbation is injected, the output
noise level for all channels varies between 2.64 to 2.94 counts.
To increase the immunity of the FEE a RF capacitor between active and return
analog power lines is placed at the input of the FEE. This DM capacitor works also
as CM filter as the negative input of the FEE is connected to the signal reference
plane to simulate the effect of a CM and DM filter as the return line is connected to
ground following one of the ground configurations detailed in chapter 2. This isshown in figure 6.11.

Filter
Cable

RBX

FEE
Copper reference
plane

Figure 6.11: CM and DM filter- Analog power input line ground connection.
1 During the test, the chip works in calibration mode, that means 1 countcorrespond to 0.33 fC
206

6.3. Susceptibility to shield currents

During the test a perturbing current with a frequency between 150 kHz and 50 MHz
and different amplitudes is injected in the analogue power line of the FEE. A
perturbation on the digital part of the FEE does not induce gradual performancedegradation, but instead generates a clock loss that requires to reset the system tore-start. However, the order of magnitude of the perturbation observed that
deteriorates the digital line is much higher than the perturbation in the analog part.
Thus limiting the analysis to the analog one is a good approach to analyze the FEE
sensitivity to conducted noise. During the test, the perturbing signal-to-noise ratio
is kept above the background noise by 33 dB µ V , as it is shown in figure 6.12.

33 dB µ V

Figure 6.12: Perturbing signal to background noise ratio

6.3 Susceptibility to shield currents
For this test a sine-wave current is injected to the power cable shield to emulate theshield currents generated by far and near electromagnetic fields coupled to theshield or by high frequency ground currents flowing in the system.

6.3.1 Noise distribution
The main characteristic of this test is that the perturbation does not distributeequally across all channels. Figure 6.13 depicts the RMS value of the amplifier
output voltages for all channels when a perturbing current of 6 mA RMS (at 5 MHz
and 10 MHz) is injected. These values are compared with the output of each channel
when no perturbation is injected (reference).

207

6. RF Susceptibility Test of FEE

Figure 6.13: Noise distribution per channel for an injected current of 6 mA - Reference, 5
MHz & 10 MHz - Measured values.
Channels 1 to 6 are placed on board A and channel 7 to 12 belongs to board B, as it isshown in figure 6.8. It is clear that some areas of the FEE are more susceptible tonoise than others. The worst area is around channels 5 and 11 and the less sensitive
area is near channels 1 and 7. The origin of this problem is found in the groundconnection between the amplifiers boards and the photo-detector board. This issue
was solved after analyzing these test results by improving the signal integrity
between the amplifiers and the photo-detector.

Figure 6.14: Variation of the RMS value of the output noise versus the magnitude of
injected current at 10 MHz - Measured values.
2086.3. Susceptibility to shield currents

Figure 6.14 shows the variation of the RMS value of the amplifier output voltage
versus to the change in magnitude of the current injected at 10 MHz for channels 1,
4, 5 and 11. These channels represent the worst case (channel 5 & 11), the best case(channel 1) and a normal case (channel 4). At low current values, the output signal
is dominated by the thermal noise contribution of the amplifier, while at highcurrents, it is dominated by the injected signal in the noisy channels.
The frequency response of all these channels to a perturbing current of 6 mA RMS isshown in the figure 6.15. At low frequencies, the current amplitude is not large
enough to disturb the channels and the output signal is dominated by the thermal
noise. At high frequencies, the perturbing signal is dominant defining the frequency
response of the charge amplifier to the coupled CM noise into each channel. From
this plot, it is possible to observe the loss of gain of the amplifier at around 17 MHz
and 34 MHz. The gain lost at 34 MHz corresponds to the intrinsic behavior of theamplifier due to the sampling at that frequency. Notches in the frequency response
at 17MHz and also at 8.5MHz appear due to the data organization in time domain.
Each amplifier, after integration during 1 ⁄ 34 µ s , send the data to the acquisition
system with a time separation of 4 ⁄ 34 µ s . This process is equivalent to a decimation
by 4 of signal sampled at 34MHz, giving the zero response at 8.5MHz and 17MHz.

Figure 6.15: Variation of the RMS value of the output noise versus the frequency of theinjected current (Ip=6 mA) - Measured values.

209

6. RF Susceptibility Test of FEE

6.3.2 Transfer function
Based on the previous data, it is possible to identify the transfer function between
the output voltage and the perturbing input signal. The transfer function is definedas the ratio between the output voltage of the FEE and the injected perturbing
current. The output voltage of the FEE can be referred to as number of counts at theoutput of the ADC or it can be referred to as the equivalent noise charge (ENC) at
the input of the amplifier, both units being related by a scale factor.

ENC
T f = -----------Ip(6.7)

The transfer function quantifies the sensitivity of the amplifier to noise currents andallows estimating the noise contribution of any perturbing signal in the system. The
transfer functions of channel 4, 5 and 11 between 1 MHz and 50 MHz are shown infigure 6.16. Lower frequencies values are not shown because the output of the FEE
is dominated by thermal noise and the transfer function is poorly estimated.

Figure 6.16: Transfer function of channel 4,5 and 11 - Measured values.
The transfer function values in the range of frequencies between 150 kHz and 100
MHz can be estimated by mathematical model of the FEE frequency response. Thismodel is estimated by combining the transfer function of the QIE1 with an external
component. The curve is parameterized based on this external component andfitted to the measured values using the least square method.
1 See chapter 1 - Section 1.7
210

6.3. Susceptibility to shield currents

The QEI transfer function is represented assuming the model of a sampledintegrator in combination with a low-pass filter that represents the frequency
response of the input stage of the device. The low frequency gain of these blocks isequal to one, including the low frequency gain of the system in the externalparameter. The following equations describe the main blocks of the transfer
function.

τ
sin ⎛ ω ⋅ ⎛ ---⎞ ⎞⎝⎝ 2⎠ ⎠
QIE = -----------------------------τω ⋅ ⎛ ---⎞
⎝ 2⎠

1
Filter = ------------------------j⋅ω
1 + ⎛ ----------⎞⎝ ωb ⎠

(6.8)

(6.9)

External parameter = ( j ⋅ ω ⋅ L m )(6.10)

τsin ω ⋅ --2
1
TF = ------------------- • ------------------------ • j ⋅ ω ⋅ Lmτj⋅ωω ⋅ --1 + ⎛ ----------⎞⎝ ωb ⎠
2(6.11)

Where τ represents the sampling period of the QIE (1/34 MHz), ω b the cut-off
frequency of the input stage (~70 MHz) and L m the parameter of the model. The
last value depends on the channel and the ground layout as described in the next
section. As an example, the measured values of channel 5 (in blue) and the fitted
function (in red) are shown in figure 6.17.

Figure 6.17: Channel 5 transfer function - Measured and fitted values.

211

6. RF Susceptibility Test of FEE

If the previous plot is compared with the ideal behavior of the QEI with a filter asshown in figure 6.18, it is clear that the FEE presents a poor CM rejection at a highfrequency, because above the sampling frequency there is a contribution of the noise
to the output of the FEE that decreases the good performance of the FEE.

Figure 6.18: QIE with filter block frequency response - Simulated values

6.3.3 Grounding configurations.
The shields of the power cables are connected to the RBX through a metallic box,
which encloses the power filter boards. The shield is connected to the metallic box
by the EMC gasket included in the cable strain relief and the box is grounded locallyto the RBX through a short metallic strap. The RBX is connected to the HCAL
ground, which is simulated during the test by a copper plane, through a strap. Thislayout is shown in figure 6.19.

Cable

Filter

Strap − Shield / RBX
Strap − RBX / REF

FEE

RBX
Copper plane

Figure 6.19: Ground connection layout between power cable, RBX and copper plane.

2126.3. Susceptibility to shield currents

During the test, the strap connection between the filter box and the RBX (marked in
red in figure 6.19) is changed to study their influence on the FEE performance. Thisconnection is made with a copper strap, 15 cm long. The length and the routing of
the strap connection to the RBX are changed. This modification produces a variationon the inductance of the ground connection. The results of three different layouts
are presented.
• GND 1: The first layout is shown in fig. 6.20a. The ground connection is donewith a long strap. It is routed to the connection point as far as possible from themetallic structure of the RBX.
• GND 2: The second layout, the strap is routed to the connection point as close as
possible to the metallic structure of the RBX following the shortest path to thatpoint (figure 6.20b without copper tape)• GND 3: The third layout is similar to the second one, but a copper tape is used to
fix the strap to the RBX (figure 6.20b). This layout decreases the length of thestrap to a minimum.

ab

GND-connection

Figure 6.20: Ground shield connection layouts.
The results previously presented correspond to the second set-up (GND 2), in
which the strap is routed as close as possible to the RBX to the ground connection,
whereas the test results for the first configuration (GND 1), which is shown in figure
6.20a are shown in figures 6.21 and 6.22. In the first plot, the frequency response ofsome sensitive channels to a perturbing current of 6 mA RMS is depicted, while
figure 6.22 shows the transfer function for several channels of the FEE.

213

6. RF Susceptibility Test of FEE

Figure 6.21: Variation of the RMS value of the output noise respect to the frequency of theinjected current (Ip=6 mA) - Measured values - GND 1 configuration.

Figure 6.22: Transfer function of channel 4,5 and 11 - Measured values - GND 1
configuration.
Figure 6.23 shows the measured values and fitted curve for channel 5. Qualitatively,
similar results have been obtained with the third layout.

214

6.3. Susceptibility to shield currents

Figure 6.23: Channel 5 transfer function - Measured and fitted values - GND 1
configuration.
The results are very similar to the test results obtained for the GND2. Identicalcomments can be applied for this test and the fitted results of the measured values.
To compare the effect of the strap connection between the power cable shield andRBX, the fitted curves of channel 5 for the three different configurations areanalyzed.

Figure 6.24: Channel 5 transfer function - Fitted values - GND 1, GND 2 and GND 3
configurations.

215

6. RF Susceptibility Test of FEE

Figure 6.24 shows the transfer function of channel 5 of the three configurations
under study. The first configuration correspond to the GND 1 ( L m = 10 pH ) , thesecond correspond to the GND2 ( L m = 7.60 pH ) and the third one correspond to the
GND3 ( L m = 5.5 pH ) .
Based on these curves, the third layout (GND 3) is the best configuration to makethe ground connection of the shield. Essentially this connection is characterized by
the shortest and less inductive strap and produces the lowest value of ground
impedance connection for the frequency range1 of interest. The system presents ahigher rejection to shield currents because most of these currents can be by-passed
from the RBX and hence they do not pass through the sensitive part of the FEE.
The ground layout represents an important issue because it defines the sensitivityof the FEE to noise coupled through the shields currents. These currents are
generated by EM fields (near and far fields) as well as by HF ground currents. Basedon this study, it is possible to define the main characteristics that should be followed
for the ground connections:• The straps should be short and flat.
• Routing path should be as close as possible to the metallic box.

6.4

Susceptibility to CM power currents

To study the effect of the common mode noise currents flowing through the central
conductors of the power cable, a perturbing current is injected to both the active
and return conductors. In practice, this CM noise is generated by the power
supplies.

6.4.1 Noise distribution
The perturbation does not distribute equally across all channels. Figure 6.25 depicts
the RMS value of the amplifier output voltage of all channels when a perturbing
current of 15 mA RMS (at 5 MHz and 10 MHz) is injected. These values arecompared with the output of each channel when no perturbation is injected(reference).

1 Chapter 2216

6.4. Susceptibility to CM power currents

Figure 6.25: Noise distribution per channel for a injected current of 15 mA - Reference, 5
MHz & 10 MHz - Measured values.
The noise distribution is the same of the previous test, being most sensitive
channels 4,5 and 11. However the magnitude of current applied to disturb the FEE
is about three times bigger. Although it will be analyzed later, the FEE is lesssusceptible to common mode currents flowing through the active and return wiresof the power cable than through the shield.

Figure 6.26: Variation of the RMS value of the output noise respect to the magnitude of
injected current at 10 MHz - Measured values.

217

6. RF Susceptibility Test of FEE

Figure 6.26 shows the variation of the RMS value of the amplifier output voltage
versus the change in magnitude of the current injected at 10 MHz for channels 1,4,5
and 11. Figure 6.27 shows the frequency response of these channels to an injectedcurrent of 15 mA.

Figure 6.27: Variation of the RMS value of the output noise respect to the frequency of theinjected current (Ip=15 mA) - Measured values
Figures 6.26 and 6.27 show that for low injection currents (at fix frequencies) and
low frequencies (at a fix amplitude) the output signal is dominated by the thermal
noise. However for high currents and high frequencies the perturbing signal isdominant. This analysis is similar to the previous test and identical conclusions canbe extended to the behavior of the FEE perturbed by common mode currents
flowing through the power cables.

6.4.2 Transfer function
Based on the previous data, it is possible to identify the transfer function between
the output voltage and the common mode current flowing through both the active
and return wires of the power cable. Figure 6.28 shows the transfer function ofchannels 4, 5 and 11 for a frequency range between 500 kHz and 40 MHz. As it was
mentioned in the previous test, the estimation of the transfer function at lower
frequencies is imprecise due to the signal to noise ratio being lower than one. The
fitted values of transfer function for channel 5 are shown in figure 6.29.

218

6.4. Susceptibility to CM power currents

Figure 6.28: Transfer function of channel 4,5 and 11 - Measured values.

Figure 6.29: Channel 5 transfer function - Measured and fitted values.
Comparing the transfer function depicted in figure 6.29 with the transfer function ofchannel 5 to shield currents (figure 6.17), it is clear that the FEE is more susceptible
to CM currents flowing through the shield than CM currents flowing through the219

6. RF Susceptibility Test of FEE

central wires. The ability of the system to filter these CM currents defines thesusceptibility of the FEE. For current flowing through the power cable shield (figure6.30), the only path to deviate the noise currents from the FEE is via the strap. For
the case of CM currents flowing through the central conductors of the power cable,
these can be bypassed from the FEE through both the shield and strap (figure 6.31).

FilterI CM

FEEI pertVout

Cable
I Shield

I strap

I stray 1

RBX
I box

I stray 2

Copper plane
Figure 6.30: Shield current distribution at the input FEE.

I CM

FilterFEE
I pertVout

Cable
I Shield

I strap

I stray 1

RBX
I box

I stray 2

Copper plane
Figure 6.31: CM current distribution at the input of the FEE.

6.4.3 Common mode filter effects.
The CM filter1 at the input power terminals protects the FEE from CM noise flowing
through the power supply lines. It has also impact in the specification of the
1 It is shown in figure 6.11
220

6.4. Susceptibility to CM power currents

common mode emission level of the power supplies. The transfer function forchannel 5 without filter at the input power terminals is calculated for better
understanding of the filter’s influence on the susceptibility of the FEE. Figure 6.32
plots the measured and fitted transfer functions for channel 5 without filter, while infigure 6.33 it is compared with the transfer function of channel 5 with filter.

Figure 6.32: Channel 5 transfer function - Measured and fitted values - FEE without input
filter.

Figure 6.33: Channel 5 transfer function - Fitted values - FEE with and without input
filter.

221

6. RF Susceptibility Test of FEE

The FEE without CM filter is about 5 times more sensitive than the FEE with filter. It
is clear that it is necessary to protect the FEE from the common mode noise currents
by avoiding these currents can flow through the sensitive FEE path, deteriorating
the performance of the system. The level of the attenuation of the CM filter showedin this analysis is relatively low, as the filter does not include any magnetic
component to improve the attenuation to CM currents. In general in the CMS
detector, the FEE can not use components with magnetic material due to the FEE
operates under strong continuous magnetic fields.
As a conclusion, to protect the FEE from CM currents flowing through the power
cables, the solution has to be addressed following two options; one is to use
common mode filters at the input of the FEE and the other is limiting the noise
emission of the power supplies to very low levels. In CMS the CM immunity of the
FEE electronics must be improved by selecting a good cabling and imposing limits
on the CM noise emission of power supplies. Power supplies with very low
common mode emission can be found in linear power supplies and switching
power supplies (SMPS) with special EMI filters. The first one presents poor
efficiency, while SMPS requiring CM emission levels too low cannot be found in themarket or they can be expensive due to they are custom designed. A solution
balancing these two strategies could be the more appropriated for these cases. A
simple CM filter built around RF capacitors located at the input of the power supply
cables combined with twisted and shield cables will provide not only good CM
rejection to CM current flowing through the central conductors but also will
improve the rejection to currents induced by far and near EM field. In addition,requiring power supplies with low CM emission will help to achieve theperformance goal of the FEE and will reduce the EM radiation from the distribution
cables.

6.5 Susceptibility to DM power currents
To study the effect of differential mode noise currents flowing through the power
cables, the perturbing current is injected into the active power cable. In addition, todecrease the CM current of the system layout, the ground connection of the power
supply and normalized common impedance box is disconnected1. In this case, theinjected current in the active power cable is forced to return through the return
power cable.
The measurements on this configuration present two constraints, which limits thevalues that can be measured. First, the layout is very sensitive to the strap’s
resonant frequency that connects the RBX to the copper plane2. This resonant
1 See figure 6.3
2 See chapter 2222

6.5. Susceptibility to DM power currents

frequency disturbs a lot the measurement, resulting the values measured to fix thetransfer function of the differential mode noise being invalid.
The second problem is that at high frequency the stray capacitance associated withthe system1 starts to play an important role increasing the value of I gnd . Therefore
the rejection to the common mode noise of the configuration decreases. Bothproblems limit the frequency range of the measurement. The box and injectedcurrent at several frequencies are measured to fix one of these limits. The results areshown in table 6.1 and 6.2.

At 5 MHzdB µ VdB µ A

mA

I DM

92.24

78.7

8.6

I strap

52.58

39.04

0.083

I box

66.01

52.47

0.42

Table 6.1 DM current distribution at 5 MHz.

At 10 MHzdB µ VdB µ A

mA

I DM

92.2

78.51

8.6

I strap

69.05

55.36

0.63

I box

75.33

61.64

1.2

Table 6.2 DM current distribution at 10 MHz.
At 5 MHz the amount of CM current that flows in the system is around 4% of thetotal current injected (differential mode around 96%), however at 10 MHz thisnumber changes to 13.4%. Therefore the DM noise that is going to disturb the FEE
maybe considered valid up to this frequency range.
Figure 6.34 shows the transfer function of channel 5, where the estimated function isfitted with values measured at lower frequencies than 10 MHz.

1 See figure 6.3

223

6. RF Susceptibility Test of FEE

Figure 6.34: Channel 5 transfer function - Measured and fitted values.
Also, this plot shows two big peaks around 18 MHz and 21 MHz, which correspondto the resonance frequencies of the strap connection between the RBX and thecopper plane. These resonances in this test set-up appear because the normalizedcommon impedance box is disconnected from ground and it does not damp theresonance as in the previous set-ups. This resonance can be moved to higher
frequencies using multi-strap connections between the RBX and the ground plane
to reduce the parasitic inductance.

6.6 DM and CM noise contributions to FEE
outputThis main goal is to analyze the effect of currents, combining common mode anddifferential mode components, on the FEE immunity and also to define normalized
specifications for the output noise of the power supplies.
The main goal of this analysis is to find out the main responsible of the degradationof the FEE performance (CM or DM noise in the central conductors) and also todefine the normalized specifications for the output noise of the power supplies.

2246.6. DM and CM noise contributions to FEE output

6.6.1 Noise distribution and TF of CM + DM currents.
The first step of this analysis is to determine the transfer function between the noiseat the input positive power lead and the output of the FEE. During this test a
perturbation current is injected through the active wire of the power cable.
The noise distribution is similar to the previous test. Figure 6.35 and 6.36 show theRMS value of the several channels versus the change in magnitude of the injecting
current at 10 MHz and versus the change in frequency for a 15 mA injecting current

Figure 6.35: Variation of the RMS value of the output noise versus the magnitude of
injected current at 10 MHz - Measured values.

Figure 6.36: Variation of the RMS value of the output noise versus the frequency of theinjected current (Ip=15 mA) - Measured values.

225

6. RF Susceptibility Test of FEE

Similar to previous analysis, it is possible to get the transfer function between the
output voltage of the FEE and the noise currents flowing through the cables. Figure
6.37 shows the transfer function of channels 4,5 and 11 for a frequency range
between 500 kHz and 40 MHz, while figure 6.38 depicts the measured and fitted
transfer function for channel 5.

Figure 6.37: Transfer function of channel 4,5 and 11 - Measured values.

Figure 6.38: Channel 5 transfer function - Measured and fitted values.

226

6.6. DM and CM noise contributions to FEE output

6.6.2 External noise contributions to the overall FEE noise
To analyze the effect of currents, with CM and DM components, on the FEE
immunity we compare this measures with the previous one where the CM and DM
components where injected independently. The current in the active line I 1 andthe CM current I cm , depicted in figure 6.39, are measured at 10MHz. Based on thesevalues, the DM current magnitude may be calculated as the difference between theactive current and CM current amplitudes. The evaluation of the DM noise current
is possible because the CM current and DM current are in phase. This is because
both DM and CM circuit present similar impedances at the input of the FEE. All thecurrent magnitudes mentioned above are shown in table 6.3.

FEE

Filter

I1 = I CM + I DM

Vout

I pertCable
I 2 = I CM − I DM

I Shield

I strap

I stray1

RBX
I stray 2

I box

HCAL GroundFigure 6.39: Layout of positive input lead immunity test of HCAL-FEE.

At 10 MHzdB µ VmA% - CM/DM

I1

96.1

13.2

100

I' cm

93.57

9.862

-

I cm = I' cm ⁄ 2

-

4.931

37

I dm = I 1 – I cm

-

8.269

63

I box

77.78

1.601

-

Table 6.3 Current distribution at 10 MHz - DM+CM
immunity test.

227

6. RF Susceptibility Test of FEE

Figure 6.40 shows the transfer function of CM, DM and positive lead modes. Basedon this plot, it is possible to define two important properties of the configuration of
the FEE:• CM currents produce more noise at the FEE output than DM currents.
• It seems that the main responsible for the degradation of the system, when acurrent is injected in the positive lead, is the CM noise flowing through thesystem.

Figure 6.40: CM, DM and CM+DM Transfer function of channel 5.
The final contribution due to a noise current flowing through a power conductor
depends on the magnitude of the CM and the DM components defined by the
impedances of the circuit and the noise transfer function for each current mode. To
verify that, a perturbing signal of 19.5 mA at 10MHZ is injected in the active power
conductors giving an output noise at channel 5 equal to 3.81 counts. This value iscompared with the noise estimation obtained by decomposing the current in its CM
and DM components and applying the CM and DM transfer functions previously
measured.
The current injected is decomposed in the two orthogonal modes, defined mainly,
by the impedance of the RBX. The common mode component of this current isequal to 7.3 mA (37%), while the differential mode component is 12.2 mA (63%).
The proportion between CM and DM components is more or less the same fordifferent current amplitudes and different frequencies up to 20 MHz.
228

6.6. DM and CM noise contributions to FEE output

Based on the CM and DM transfer function at 10 MHz previously calculated, thenoise at the output of channel 5 generated by these currents are:fC
countsCounts – ch 5 CM = 0.048 -------- × 3 ----------------- × ( 7.3 × 2 ) mA = 2.101 counts
mA
fC
fC
countsCounts – ch 5 DM = 0.023 -------- × 3 ----------------- × 12.2 mA = 0.841 counts
mA
fC(6.12)(6.13)The total numbers of the counts at the output of channel 5, including the thermal
noise of the amplifier is:

Ch 5 =

22( 2.101 + 0.841 ) + ( 2.91 ) =2

2( 2.942 ) + ( 2.91 ) = 4.1counts(6.14)

This value is in close agreement with the measured value, which is 3.81 counts. The
error is around 7.5%. The first term of the square root (2.942) defines thecontribution of the external noise to the total FEE output noise. Based on thisnumber, it is possible to mention that the 71% of the contribution of the external
noise is generated by CM currents and the 29% has its origin in DM currents, when
the DM current is 1.7 times larger than the CM current. So, it is possible to conclude
that the main responsible of the degradation of the FEE noise immunity are the CM
currents.

6.6.3 Normalized noise level required at the output of the PS
Based on the transfer functions estimated when CM or DM current is injected, theeffect on the FEE output noise has been analyzed when the sensitive electronics is
powered up by switching mode power supplies (SMPS). Although, only a few
standards define the conductive noise level at the power supply output, the noiselevel required at SMPS outputs for optimal performance of the FEE should beestablished.
Using the HCAL sub-system as reference, the effects of conducted emissions forthree different FEE topologies have been studied to identify the standardized noiselevel required at the power supply output. The topologies used during this studies,correspond to a FEE with good EMI configuration (A), a FEE without EMI input
filter (B), and a FEE with a bad EMI design, (C). Theses configurations arecharacterized by the parameter value (Lm) used in this chapter to parameterize thetransfer functions. The values of these parameters for conducted emissions on thepositive terminal are 3 pH for configuration A, 15 pH for configuration B and 75 pH
for configuration C and for the CM emissions are 2.7 pH for configuration A, 13.5
pH for configuration B and 67.5 pH for configuration C. Figure 6.41a shows thetransfer functions of the conducted noise at the input positive terminal of the three229

6. RF Susceptibility Test of FEE

configurations of the FEE and figure 6.41b shows the transfer function of CM at the
input of the FEE of the three configurations of the FEE.

ab

Figure 6.41: a) TF of positive input current b) TF of CM input current for A,B and C FEE
configurations.
The conducted emissions assumed correspond to three different theoretical power
supply units whose emissions are equal to the level defined by the standards EN
55022 class A or class B, and the switching frequency of each PS units is set todifferent magnitudes as; 100 kHz, 200 kHz and 300 kHz. Figure 6.42a shows theoutput voltage spectrum at the positive terminal of the PS switching at 200 kHz
complying with the EN 55022 class A standard, while figure 6.42b shows the CM
emission of this PS following the level defined by CM-EN 55022 - class A.

ab

Figure 6.42: Conducted noise emissions of a theoretical PS units switching at 200 kHz.

230

6.6. DM and CM noise contributions to FEE output

EN 55022 - A

A

B

C

EN 55022 - B

A

B

C

100 kHz

2.91

3.12

6.41

100 kHz

2.91

2.92

3.17

200 kHz

2.91

3.02

4.97

-

2.91

2.91

3.04

300 kHz

2.91

2.98

4.39

-

2.91

2.91

3.00

Table 6.4 Estimated output noise for channel 5
generated by conducted emissions at the positive
terminal of the FEE.

CM / EN 55022 - A

A

B

C

CM / EN 55022 - B

A

B

C

100 kHz

2.92

3.33

8.64

100 kHz

2.91

2.95

3.43

200 kHz

2.91

3.12

6.44

-

2.91

2.92

3.18

300 kHz

2.91

3.05

5.51

-

2.91

2.91

3.09

Table 6.5 Estimated values of FEE channel 5 generatedby CM emissions at the input terminal of the FEE.
Tables 6.4 and 6.5 summarize the estimated output noise for channel 5 for different
conducted emissions and different FEE configurations. These values have been
calculated using the DM+CM TF and CM TF previously defined in figure 6.41. FEE
units characterized by a high noise immunity show good performance when they
are biased by power supplies whose conducted output noise comply with EN55022
class A. In opposition, FEE units with poor noise rejection require power supplies
with very low noise output voltage. These power supplies have either to comply
with military or FCC - Class B standards or they have to be linear power supplies.
Similarly, table 6.5 shows the same tendency for power supplies having dominant
CM output noise emission. This table shows that the combination of high CM
emission at the power supply and poor CM rejection at the FEE gives a poor overall
performance even in the case that no DM noise emission or ripple exists. It shows
how important is to define the maximum level of emission for CM currents.
For the previous analysis it has been considered that power supplies emit noise inwhole frequency range defined by the standards. In general, DM emissions are
more important at low frequencies whereas the CM is more important above theMHz range. Also, analyzing more in detail the immunity curves, it is clear that the
FEE is more susceptible to noise at high frequency than at low frequency and thenoise contribution extends above 30MHz that is the maximum frequency set by the231

6. RF Susceptibility Test of FEE

EN 550222 standard for conductive noise. It seems that the noise level fixed by theEuropean standards do not define the optimal requirements for the HCAL
sub-system, being necessary to define a new emission level that adapts better to thenoise sensitivity of this FEE.
The definition of the noise level will be based on the inverse curve of the transfer
function that defines the immunity level of the FEE. The noise level required at theoutput of the power supply is defined by a constant curve above 20 MHz and below
of this frequency, the level increase at 20 dB per decade up to 100 kHz. To fix thelevel of the proposed limit curve, a new noise analysis is carried out. The effect on
the output noise of the FEE biased by a PS with an output conducted emission as
defined by the proposed limit is calculated. It is assumed the power supply
switches at 100kHz and the frequency range for the noise spectrum is extended
between 100 kHz to 50 MHz, because there is an important noise contribution at theoutput of the FEE above 34 MHz. A criteria to define the level of the proposed limit
can be fixed when the output noise of channel 5 is equal to 3 counts for the
conduction emission in the positive terminal (noise contribution equal to 25% of thethermal noise) and 2.97 counts to define the level for the CM noise (20% of thethermal noise). Following these criteria the level depends on the characteristic of the
FEE being necessary to establish a curve for each FEE configuration. Figure 6.43
shows the levels of the proposed limit corresponding to the different FEE
configurations and compare those limits with the European standards. Figure 6.43a
depicts the limits for the terminal current, while figure 6.43b shows the limits for the
CM currents.

ab

Figure 6.43: European (EN 55022 - Class A & B) and HCAL standard levels for positive
input teminal noise and CM input noise.

232

6.6. DM and CM noise contributions to FEE output

In general, this plot shows a big difference between the requirements for the HCAL
FEE and the levels imposed by the standards. For good EMI configurations the level
required at the output are much more relaxed than the European norms. However,if bad configurations of the FEE are designed the level required are very restrictive,
being for certain frequency range extremely low. Also, there is an importantdifference respect to the European standards at high frequency. Due to the FEE issensitive above 30 MHz it is necessary to extend the emission frequency spectrum
considered up to 50 MHz. This is a particular consideration for the HCAL FEE, forother sub-systems the spectrum considered could be extended to higher frequencies
or not depending upon the FEE sensitivity. The FEE output noise (total number ofcounts of channel 5) for different power supplies switching at 100 kHz with output
noise characteristics level equal to each of the standards levels defined above anddifferent FEE configurations (A,B and C) are summarized in the table 6.6. The basenoise for this channel is 2.91 counts.

DM+CM

A

B

C

CM

A

B

C

EN 55022 - A

2.91

3.12

6.41

CM EN 55022 - A

2.92

3.33

8.64

EN 55022 - B

2.91

2.92

3.17

CM EN 55022 - B

2.91

2.93

3.43

HCAL - A

3.03

5.69

24.6

CM HCAL - A

2.97

4.25

15.81

HCAL -B

2.91

3.01

5.23

CM HCAL -B

2.91

2.98

4.53

HCAL - C

2.91

2.91

3.03

CM HCAL - C

2.91

2.91

2.97

Table 6.6 Estimated values of FEE channel 5 generated
by conducted emissions at the positive terminal and CM
emissions at the input FEE.
As an example of the application of this new standards levels the conductedemission of a VICOR DC-DC converter is evaluated. Figure 6.44a shows the
conducted emission at the output of the Vicor converter without EMI filter andfigure 6.44b shows the CM emissions of the same unit. The noise emission of the
VICOR converter complies with the emission level defined by HCAL-A. When thisconverter bias the FEE HCAL-A, the noise output of channel 5 is dominated by thethermal noise (Ch5-A = 2.91 counts) and the FEE performance is not affected.
However, this emission level is quite far to comply with any of the European
standards. A similar analysis is valid for the CM. The noise level of this Vicor
converter is close to comply with the curve level defined by HCAL-B. For
frequencies in the low-range of the spectrum, the output noise is large than thelimit, while for higher frequencies, the noise lower, given this combined effect anoverall noise at the FEE output of 3.05 counts. This converter does not comply withthe level defined by HCAL-C, given a total noise of 5.28 counts.

233

6. RF Susceptibility Test of FEEab

Figure 6.44: Conducted and CM emissions of Vicor converter compared with different
standardized levels.
If the Vicor converter includes an output filter, the emission level is depicted infigure 6.45. For this case, the output noise emission complies with both the HCAL-A
and HCAL-B levels, while at some frequencies it does not comply with the HCAL-C
level. In this last case, if the FEE HCAL-C is biased by this converter with filter the
output noise is equal to 2.96 count, being an acceptable performance.

ab

Figure 6.45: Conducted and CM emissions of Vicor converter with EMI output filter
compared with different standardized levels.

234

6.7. Summary

It is necessary to remark that none of these emissions fits the requirements fixed by
European standards, however even FEE with bad EMI configurations might workwithout any problem. It is very important to define the immunity level of the FEE
for the different sub-systems. Each of them will be different and the application of
the standards without the knowledge of this level can lead to serious problems.
However, if immunity level curves are defined, it is possible to set a new level ordefine the European standards level that fits better to the requirements of thedetector. For the present example the selected level would be Class A for good EMI
immunity and Class B for FEE without very good EMI immunity. For the case ofvery bad EMI topologies, it would be necessary to apply more restrictive levels like
FCC class B or military one. In all cases the standard level would be extended infrequency up to at least to 50 MHz.

6.7 Summary
The susceptibility of the HCAL FEE to noise currents in LV power supply inputs isdefined by the susceptibility of the system to CM currents, DM currents and shieldcurrents. The susceptibility is quantified by the transfer function between the
output of the FEE and the perturbing current.
The principal cause of the degradation of the FEE immunity is CM currents. These
CM currents can be coupled into the FEE by shield currents, by CM flowing
through the cables and by the DM to CM conversion. The DM mainly contributes tothe degradation of the FEE because of DM to CM conversion.
There are several actions that can be applied to improve the FEE immunity to noise
currents.
• Select good ground connections• Use CM filters• Use of shielded power cables
The first one is defined by the low impedance of the ground connections in thewhole frequency range, from a few Hz up to several tens of MHz. It is
recommended to used very short and multiple straps to make the groundconnections as well as to take care to route theses ground connections as close as
possible to the ground or reference plane. These recommendations are focused increating a low impedance-grounding path for the system, which helps to bypass thenoise currents from the sensitive parts of the detector. Also, it is recommended to
weld these connections or use very tight connections which ensure a very good
contact to ground of the FEE.
The second recommendation is focused on filtering the common mode noise
flowing through the conductors. This filter must be implemented with RF235

6. RF Susceptibility Test of FEE

capacitors because the high magnetic field at the FEE area does not allow the use ofmagnetic chokes. These simple filters are not as good as a real EMI filter, however
they improve the performance of the system in presence of common mode noise.
This solution used in combination with the limitation to low levels of the common
mode noise generated by the power supplies, which can be easily achieved,complete the implementation of a robust system with low noise output voltage.
The final recommendation is oriented to avoid the noise radiation through thecables and to give a low impedance path to CM noise currents to avoid thementering into the FEE. The use of un-shielded power cables, should not be used, as it
will force the selection of power supplies with very low CM emission levels at theoutput, which it is not practical for technical and budgetary reasons.
The immunity curves of the system can be used to define a new noise level requiredat the output of the power supply. For HCAL FEE this level results very different
from the level defined by the European standards. The definition of a standardized
noise level at the output of the power supplies without the knowledge of theimmunity curves of the system can lead to problems. It is important the definitionof this level for all the sub-systems of the detector. Later, based on this curves, it ispossible to define the emission level for the power supplies or select the European,
American or Military standards that fits better to the requirements of the detector.
In any case, the standardized level should be extended above the 30MHz limit
because the FEE is very sensitivity in that frequency range.

6.8 References[1] Clayton R. Paul, "Introduction to Electromagnetic Compatibility", NY Wiley
Interscience, 1992,ISBN-0-471-54927-4.
[2] T. Williams, "EMC for Product Designers", 1997, ISBN-0-7506-2466-3.
[3] F. Arteche, C. Rivetta and F. Szonsco "Electromagnetic compatibility plan for the
CMS detector at CERN ", Proc. 15th International Zurich Symposium and Technical
Exhibition on Electromagnetic Compatibility, Zurich, Switzerland, pp. 533-538,
February 2003.
[4] International Standard IEC-61000-4-6, "Electromagnetic compatibility (EMC)
Testing and measurement techniques - Immunity to conducted disturbances, induced byradio-frequency fields", Basic EMC publication (Ed. 2000).
[5] T. Shaw et. al., "Front End Readout Electronics for the CMS Hadron Calorimeter",Proc. IEEE Nuclear Science Conference 2002, Viginia, USA, Vol 1, pp 194-197,
November 2002.
[6] Dan Green, "HCAL Status", Oral presentation, CMS Week June 2003.

236

Chapter 7
Susceptibility to transient
perturbation

This chapter addresses the analysis of transients in the power supply system of
CMS. Transients in power distribution lines can be generated by variations of thecurrent consumption of the FEE due to changes in operation mode, random failures,
equipment turn on/off, etc. They can induce over-voltage and voltage dips in the
LV-DC and AC distribution system, which can affect the performance and thereliability of the front-end electronics. During this analysis the CMS power supply
distribution is divided in two areas. The first one corresponds to the AC
distribution (between the counting room and the periphery of the detector) and theother to the DC distribution (between the periphery and the FEE). The transient
limits for the first area are well defined by the standards, this is a typical AC
distribution. However, limits for the second one are not clear and they depend on
the characterization of the FEE to this kind of transients, power supply distribution
topology, cost and risk. This chapter is focused mainly on this part. The main goal is
to describe possible sources of transients and the implications that they have on the
FEE performance. The chapter does not present neither the final transients levels of
CMS nor the voltage level to be applied to the FEE during the transientssusceptibility test because these values depend on the final topology of the power
supply distribution system of CMS, which has not been defined yet.

7.1 Introduction
The low and high voltage distribution system in CMS is based on a primary 400Hz
AC system. As it was described in chapter 2, it is composed of independent
motor-generator sets for each sub-detector that converts the 50Hz AC mains to a
400Hz three-phase 400V system. The AC 400Hz-distribution system feeds AC/DC
converters located around the detector to power up sub-detectors. These units are
simply three phase step-down transformers with rectifiers and LC filters in thesecondary. They convert the AC high voltage into appropriate DC low voltages that237

7. Susceptibility to transient perturbation

are locally distributed around the detector. In this case, local LV regulation is
performed in the front-end electronics at the board level using radiation tolerant
low voltage drop regulators. Other sub-systems will convert the AC to 48V DC anddistribute this intermediate voltage to DC-DC switching converters that supply the
final power to the FEE. The HV power supplies are located in remote areas awayfrom the detector as high energy particle radiation can induce on these systems
catastrophic failures. In general, this voltage distribution involves cables withlength larger than 100 meters.
The environmental condition at the detector is full of transient interactions. Fast andslow transients and voltage dips, can be found in the power supply distribution.
Transients can induce EM interactions that affect the performance or reliability of
the front-end electronics. The order of magnitude of the transient as well as theduration is a design parameter that is defined by the configuration of the power
supply distribution and the susceptibility of the FEE. It has serious impact on theselection of the system protection (rates, reaction time etc.). The design of thesystem [1] should be focused to allow higher transients level for short time periods.
Figure 7.1 shows the design goals of a typical power supply distribution system.

Figure 7.1: Design goal for a power supply distribution system fix by IEEE-Std 1100-1992.
Surges or slow transients are generated by switching transients or by lightning
strokes injecting high voltage/currents via electromagnetic fields. The latter are notconsidered in the present study as the detector is placed 100 m underground and iswell protected against these faults at surface level. Switching transients can begenerated by power system switching, load changes, short circuit faults to theearthing system of the installation. The slow transient voltages are characterized bya fast rise time (a few µ s ) followed by a decay time of some tenths or hundredths of238

7.1. Introductionµ s . The energy involved in these transients is high and can cause the destruction of
the electronic devices in the detector.
Fast transients are generated by switching transients (interruption of inductive
loads, relay contact bounce, etc.). The fast transient is characterized by a very fastrise time (a few ns ) followed by a decay time of some tenths of ns . In some cases,
if the switching arc is unstable, as the contacts open, the current is often interrupted
and re-ignited several times (bouncing) before the circuit is in open stable condition.
The energy involved in these transients is not very high (a few mJ) but can cause
malfunctions of the electronics devices in the detector.
Voltage dips and short interruptions are caused by faults in the network or in the
installations, by a sudden large change of load, by inrush currents energizing
transformers, etc. Voltage variations are caused by the continuously varying loads
connected to the network. The duration of these transients are between a few ms up
to several tenths of ms.
Since these disturbances are unavoidable in a distribution system with the power
level required in CMS, it is necessary to include a combination of protection devices
in the design of the sensitive electronic equipment. The potential impact oftransients depends on its severity and the susceptibility of the equipment. Threetypes of occurrences are possible:• Signal-data disruption: Circuits carrying signals are susceptible to surgeinterference via conductive, inductive, capacitive and electromagnetic coupling.
They can affect the integrity of the signal.
• Gradual hardware stress and latent failures.
• Immediate hardware destruction.
This chapter addresses the study of transients in CMS. The generation, thepropagation, order of magnitude and duration of the transient are studied as well as
the solutions that can be used to improve the immunity of the electronics against
these phenomena. These studies are based on Pspice simulations and somemeasurements on FEE prototypes of HCAL and tracker sub-systems. These analyses areused to obtain general rules, which help to tackle the transient problem in an earlystage of the design and take the corrective actions, which ensures the immunity of
the FEE against transients.

239

7. Susceptibility to transient perturbation

7.2 Voltage dips and short interruptions
7.2.1 Introduction
All electrical and electronic devices can be affected by voltage dips or shortinterruptions of power supply [2]. A voltage dip is defined as a sudden reduction of
the voltage during a short period of time, from a few ms to a few seconds, at a pointin the electrical system followed by a voltage recovery. Short interruption is definedas the disappearance of the supply voltage for a short period of time. It may beconsidered as voltage dips with 100% of amplitude. Typical waveforms of thesetransients are shown in figure 7.2. These transients are generated by faults in the
network, by a sudden large change of load, by in-rush currents, etc. [3][4].

Figure 7.2: Voltage dips standards wave form and magnitude.
These phenomena are random in nature and may be characterized in terms ofdeviation from the rated voltage and duration. Voltage dips and short interruptions
are not always abrupt, because of the reaction time of rotating machines andprotection elements connected to the power supply network.
The step load changes are one of the most common sources of voltage disturbance.
The basic cause of the voltage disturbance is simply the voltage drop caused by theabrupt load current change and the power system impedance. The inrush currents
energizing transformers and filter capacitors in power supplies can reach values ofabout 10-20 times the nominal current if no limiting circuits are included in theequipment. The transformer start-up creates a magnetizing current transient,
decaying to steady state after several cycles under worst-case conditions. This issue
is particularly critical for transformers located in the periphery of the detector
where the detector's fringe field pre-magnetizes the core of the transformers. In
addition, when rectifier/capacitor power supplies are energized, the initial current
charging the capacitor can cause similar levels of inrush currents. The fault currents2407.2. Voltage dips and short interruptions

represent an extreme case of transient current and thus AC or DC line voltage
disturbance. Depending on the power system impedance, several order ofmagnitude of nominal load current may be available. Severe voltage reduction in
adjacent equipment usually results until the fault is cleared.

7.2.2 HCAL FEE susceptibility test to voltage dips and shortinterruptions.
The goal of this test is to determinate the susceptibility of the equipment to shortvoltage interruptions and voltage variations of the primary power supply and tocharacterize the behavior of the LV regulators placed on the boards. Shortinterruptions and voltage variations are applied to the FEE. The degradation of the
FEE performance and the loss of acquired data are evaluated. The test uses as guidethe standard EN-61000-4-11/ IEC-1000-4-11 [5]. This test is focused on
characterizing the transient levels for the area between the periphery of the detector
and the detector itself. For the other area (AC distribution - Periphery of thedetector to counting room) both amplitude and duration of the transients are
perfectly defined by the standards and the applications of the standards are enoughto fulfill the requirements of the detector.
This test was conducted to the same FEE system as described chapter 6. Voltage
dips were applied independently to the voltage feeding each FEE power cable,using the transient generator (UCS-500) and two power supplies, which form thetwo voltages needed by the UCS-500 to create the voltage dip. The loss of data orloss of function was used as criteria to qualify the performance of the front-end
electronics. The test has been carried out in the analog and digital part of the FEE.
The test layout is shown in figure 7.3

Figure 7.3: Voltage dip test layout.
The voltage at the back-plane (input of the LV regulator - V1) and at the output of
the LV regulator (V2) have been measured to study the behavior of the LV
regulators during the voltage dips. This is shown in figure 7.4.
241

7. Susceptibility to transient perturbationV1

4 .5 V5 .5 VP C b o a rd
6 .5 V3 .3 V / 2 .5 V
L in e a r
R eg.

7 .5 V

L in e a r
R eg

5 .5 v
6 C h a n n e ls fr o n t-e n d
e le c tr o n ic s5 .5 v
L in e a r
R eg

V2
Figure 7.4: RBX layout - Measured voltages
Figure 7.5 shows the transient voltage at both the input and the output of the LV
regulator during a voltage dip. There are three operation areas of the LV regulator.
In the first one, corresponding to area A in figure 7.5, the LV regulator works
properly because the input voltage is higher than the minimum input voltage
required. The output voltage of the LV regulator is constant independent of the
input voltage. In the second area, area B, the LV regulator cannot regulate theoutput voltage and behaves as an impedance. The output voltage drops at the sametime as the input voltage. Later, in area C, when the voltage reaches a threshold thevoltage regulator drops the output voltage to zero.

A

B C

B

A

Figure 7.5: Input and output regulator voltage during a voltage dip.
Voltage dips were applied independently to both the analogue and digital power
supplies feeding the HCAL-FEE and the loss of data or loss of function was used as
criteria to qualify the performance of the system. The results of this test are shownin tables 7.1 and 7.2. The two upper rows resume the operation when the input
voltage is applied continuously without any perturbation superimposed but with a242

7.2. Voltage dips and short interruptions

magnitude lower than the nominal voltage. The rest of the table describes theoperation when the input voltage is kept at the nominal value (6.5V for analogue,
4.5V for digital) and repetitive voltage dips with different amplitudes and durations
are applied. When the voltage is applied to the analogue power supply, the only
effect detected is a simple gain loss of the amplifier, which recovers when the dip
finishes. When the voltage is applied to the digital power supply, two problems
have been found. The first one is a loss of data and the second one is a clock failure.
In case of clock failure, the FEE has to be re-setted. These faults define that thedigital part of the FEE is more critical to these transients because in case that theclock signal is lost the system cannot be recovered and it is necessary to reset the
sub-detector.
Minimum
Voltage-V1 (Volts)

Voltage Dip(Volts)Dip
duration (ms)

StatusSystemrecovery

5.15

-

Cte

OK

-

5

-

Cte

Gain loss

Yes

5.12

1.38

3

OK

-

5

1.5

3

Gain loss

Yes

4.8

1.7

3

Gain loss

Yes

Table 7.1 Results for voltage dips applied to theanalogue power supply line - (Vdip = Vnom-Vmin).

Minimum
Voltage (Volts)

Voltage Dip(Volts)Dip
duration (ms)

StatusSystem recovery

3.2

-

Cte

OK

-

3

-

Cte

Loss of Data
Clock fails

No
Reset system

3.3

2.2

5

OK

-

3

2.5

5

Loss of Data

Yes

2.3

3.2

5

Loss of Data

Yes

3.3

2.2

10

OK

-

3

2.5

10

Loss of Data

Yes

1.88

3.62

10

Loss of Data
Clock fails

No
Reset system

Table 7.2 Results for voltage dips applied to the digital
power supply line - (Vdip = Vnom-Vmin)

243

7. Susceptibility to transient perturbation

The easiest method to achieve immunity to voltage dips is to make sure that the DC
distribution has enough ’hold up’ times. This is usually achieved by using higher
values of capacitance on the DC bus bars and also by increasing the unregulated
voltage to allow more headroom for the voltage dips. It is necessary to have enoughenergy storage, given the power consumption of the product, to ride through thefrequent short dips and dropouts. If non-essential functions can be switched off
during a ’hold up’ event, or the microprocessors can be put into ’sleep mode’, it will
increase ’hold-up’ time during voltage-dip. When the dips are so long that they
become a sag or interruption and the unregulated energy storage cannot ’hold up’
any longer, it is important to ensure that before the logic DC supplies go out of
tolerance the product ’cleanly’ (no memory is lost or corrupted and no unintended
operations) and safely ceases operation and shuts down. Then, when the event has
passed, operation can be cleanly and safety restored

7.3 Surge or slow transients
7.3.1 Introduction
In the CMS detector, slow transients are generally generated by switching effects. A
few examples are listed below:• Inrush currents generated by energizing capacitors and transformers.
• Load changes in the power distribution system.
• Resonating circuits, such us long cables plus filter capacitors associated withswitching devices.
• Short circuits and faults to the earthing system of the insulation.
The slow transient waveform is characterized [4][6][7][8] by a fast rise time (a fewµ s ) followed by a decay time of some tenths or hundredths of µ s . The energy
involved in these transients is high (few mJ up to some Joules) and can cause thedestruction of the electronic devices in the detector. In the case of the CMS detector,
the victim circuit is characterized by very low impedance, which makes the abilityof the noise sources to couple to the victim (FEE) much easier.
The power supply distribution system can be divided in two areas, one the
400Hz-AC power system and the other the DC power distribution system. The
transient characteristics of each one are different, making necessary to tackle thetransient problem separately. The AC power distribution system is extended
between the surface level or counting room to the periphery of the detector (around
100 meters or 250 meters). It corresponds to the typical AC power distribution,
which is well defined by the standards [2][9]. The test to be applied for defining theimmunity of the equipment is defined by the standard IEC-61000-4-5. This standard244

7.3. Surge or slow transientsdefines 5 different levels depending on the characteristic of the electricalenvironment where the equipment is going to be installed. A brief description ofeach class follows; a well-protected environment characterizes an electrical systemclassified as Class 0. In this system all incoming cables are provided with overvoltage (primary and secondary) protection. The units of the electronic equipment
are interconnected by well-designed earthing system. Class 1 electrical system isdefined by an environment partially protected, in which the electronic equipment
has its own power supply completely separated from the other equipment andswitching operations can generate interference voltages within the room. Class 2
electrical system is characterized to have an electrical environment, where thecables are well separated even a short runs. An electrical environment where powerand signal cables run in parallel characterizes an electrical installation classified as
Class 3. This type of installation is earthed to the common earthing system of thepower installation, which can be essentially subjected to interference voltages
generated by the installation itself. Current due to earth faults, or switching
operations may generate interference voltages with relatively high amplitudes inthe earthing system. On the other hand, a Class 4 electrical environment ischaracterized to have cables for both electronic and electric circuits running inparallel. Currents of kA range due to earth faults and switching operations may
produce interference voltages of high amplitude in the earthing system of this kindof environments. For this environment the power supply network can be the samefor both electrical and electronic equipment. Also, class 4 electrical environment aresystems in which there is no systematically constructed earthing network outside
the electronic equipment, and the earthing systems consist of pipes, cables etc.
Finally, an electrical environment classified as Class 5 corresponds to an electricalenvironment where the electronic equipment is connected to telecommunication
cables and overhead lines in a non-density populated area.
According to the previous descriptions, all equipment placed in the AC area can beclassified as either class 3 or 4. In this case, positive and negative pulses ofamplitude equal to 2 kV (Line-Line) and 4 kV (Line-Ground) will be applied during
surge tests following the test procedure well defined by the standards.
The DC power supply distribution system is located between the balconies(periphery of the detector) and the detector. The transients generated by the AC line
are attenuated at the input of the DC system, being the transients generated by theDC system itself the only one that can affect the detector (FEE). This area is the mostcritical in terms of the protection selection because the high magnetic field and theradiation do not allow to use typical protection devices applied in normal
environments. The transient analysis is focused in this area of the detector. This
study is split in two parts, the former consists on the analysis of the transient effects
generated by load changes, magnetic coupling and short-circuits in the DC area of
the CMS detector. The magnitude, the duration and the energy involved in thetransient are studied. This study is based on Pspice simulations of preliminary
designs of power supply distributions to be applied in CMS sub-detectors. The

245

7. Susceptibility to transient perturbation

second part of the study is focused on the surge test. The goal of this study is toanalyze the effects and internal voltage distribution in the FEE of surge
perturbations. A surge signal of certain amplitude in DM and CM configuration ofboth positive and negative polarities is applied to a prototype of the HCAL
sub-system, and the effects on a dummy board of the system are studied.
These examples are used to describe the implication that some PS distribution
topologies have on the magnitude and duration of transients as well as the impactthat these topologies have in terms of the protection selections, risks and costs. This
analysis can be used as a base to define a transient immunity test, which allows to
do the integration of the equipment in a safe way. This section does not pretend to
give the test level to be applied to the detector because this level is defined by the
final topology of the power supply distribution, which is not known yet. However,this study is the initial point to carry similar analysis with other sub-systems todefine final tests to apply to the CMS detector electronic equipment.

7.3.2 Transients in DC distribution
The transients generated by a sudden load change, magnetic coupling and theenergy storage in a cable during a short circuit are studied in this section. Twostudies are carried out on the DC power supply distribution of the tracker
sub-detector. The Tracker sub-system is powered by around 2000 power supplies
units located 100 or 40 meters far from the detector with remote sensing to control
the voltage at the FEE power terminals. Each power supply generates two very low
voltages to power the FEE (+2.5 V, +1.25V) with a common power return and thehigh voltage (600V), which is used to bias the detector device. The front-end
electronics is characterized by a very low over-voltage margin in the bias voltage,around 200 mV, which force to control accurately any transient generated in the LV
line, such us load changes. The power supply units are connected to the detector
using a very complex power cable where LV power wires, shielded twisted pairsense wires, shielded detector bias wires and shielded temperature sensor wires areplaced together inside an aluminum shield. With this cable topology, transients
generated by load changes or magnetic coupling between lines that carry different
power levels may induce catastrophic failures in the FEE. The low over-voltage
margin of the FEE and the environment, characterized by high radiation, where thetracker sub-detector is located limit the use of classical protection devices and force
to study these transients in detail.
A final study is focused on transients generated by the energy stored in a cableduring a short circuit. A large current runs in a cable during a short circuit
energizing the cable. When the protection device opens the circuit, the stored
energy induces an over-voltage at the end of the cable (distribution bus), which may
produce serious damages to all the downstream FEE connected to the same bus.

246

7.3. Surge or slow transientsThe granularity of the DC power supply distribution system as well as the distance
between the power supply units and the FEE have a direct relation to the order ofmagnitude of these transients, producing a direct impact in the selection ofprotections devices. This study is carried out in a generic power supply distribution
system of the CMS detector. The main goal of this studies is to describe thecharacteristics of the slows transients in the DC power supply line to define the typeof the test to apply to the equipment installed in this area.
7.3.2.1 The Tracker DC power distribution system
The tracker power supply system1[10] is designed to supply power to a group of
FEE with two different voltages and currents (+ 2.5 V / 11 A and + 1.25 V / 5 A)
with a common power return. Also, these units supply the HV necessary to bias thedetector (+600 V-10 mA). Due to constraints in the distribution channels of thedetector, this long cable is divided in two sections of different lengths and wiresections. Some examples of over voltages generated by sudden load changes andmagnetic coupling in one module of the power supply distribution system of the
tracker detector are presented in this section. The frequency range of thesetransients is the order of 1 kHz-30 kHz, which allows to simplify the cable model
presented in chapter 4. Equations 4.1 and 4.2 can be simplified for the tracker cablefollowing an approximated model defined by the ’lumped’ equations:zz d
V 2 – V 1 = R ⋅ --- ⋅ I 1 ( t ) + L ⋅ --- ⋅ I 1 ( t )
2
2 dt

(7.1)

z
z d
V3 – V2 = R ⋅ --- ⋅ I 2 ( t ) + L ⋅ --- ⋅ I 2 ( t )
2
2 dt

(7.2)

I2 – I 1 = C ⋅ z ⋅

d
V (t)
dt 2

(7.3)

where L,C and R are the matrices of the characteristic parameters per unit length, z
is the length of the line and 1 is the beginning, 2 midpoint, and 3 endpoint of theline. Based on this simplification, each line can be represented in Psipce simulator.
The simulated model corresponds to one module of the tracker detector, one power
cable and one power supply unit. The module is formed by a group of FE-hybrids
with 48 APVs. The nominal current consumption per line for the group is equal to
11 A for the 2.5V line and 5 A for the 1.25V line.
The LV channel of the FEE is simplified by the FE-hybrids and two identical input
capacitors (Cb = 483 µ f ). A current source in parallel with a damping resistance isused to represent the equivalent circuit presented by the FE-hybrids at the input
power terminals. The values of these resistances are estimated by measures of the
1 The tracker power supply system is described in chapter 1.

247

7. Susceptibility to transient perturbation
dynamic impedance of 4-APVs tracker outer barrel hybrids1 [11]. The damping
resistances for the model are equal to 0.6 Ω for the 2.5V line and to 1 M Ω for the
1.25V. The HV of the FEE is represented by a resistance of 1 M Ω , which represents
the high impedance of the detector, and by the HV input filter, which is represented
by a resistance of 10 k Ω and a capacitor of 22 nF.
The LV power supply unit is formed by a battery (V1=6.4 V, V2 = 4.6 V), two output
capacitor in parallel and two switches. The two output capacitors are identical andtheir nominal values are Cps=500 µ F . The HV power supply line is represented
only by the outer capacitor, whose nominal value is 300 nf. This capacitor is charged
initially to 600 V (nominal value of the HV line).
The cable model corresponds to two sections of tracker cable (one of 40 meters andthe other of 60 meters). Only the three LV wires (2.5 V-1.25 V-LV common return)and two HV wires (+600V-HV return) have been included in the model. The
per-unit length parameters at 11 kHz of ’section 1-40 meter’ tracker power cable havebeen measured following the procedure explained in section 4.3.1. Pspiceparameters of the lumped model are calculated based on these measurements. The
values for the second section have been estimated from the values of section 1 inwhich the values of L and C have been kept constant changing only the resistance ofthe cable. All of these values are shown in table 7.3 and 7.4. The skin effect has notbeen considered in this analysis. The validation of the cable model presented isshown in appendix B.
InductanceµH-------m

L11

0.424

8.48

µH

12.72

µH

L22

0.632

12.64

µH

18.96

µH

L33

0.501

10.02

µH

15.03

µH

L44

1.188

23.76

µH

35.64

µH

L55

1.176

23.52

µH

35.28

µH

L12

0.272

k12 = 0.525

k12 = 0.525

L13

0.224

k13 = 0.486

k13 = 0.486

L14

0.225

k14 = 0.317

k14 = 0.317

Section 1 - 40 m
Pspice lump values

Section 2 - 60 m
Pspice lump values

Table 7.3 Pspice inductance values of tracker power
cable - MSC (LV and HV wires).

1 FE-hybrids curves are shown in appendix B
248

7.3. Surge or slow transients

InductanceµH-------mSection 1 - 40 m
Pspice lump values

Section 2 - 60 m Pspice
lump values

L15

0.207

k15 = 0.293

k15 = 0.293

L23

0.235

k23 = 0.417

k23 = 0.417

L24

0.328

k24 = 0.378

k24 = 0.378

L25

0.312

k25 = 0.361

k25 = 0.361

L34

0.373

k34 = 0.483

k34 = 0.483

L35

0.358

k35 = 0.466

k35 = 0.466

L45

0.668

k45 = 0.565

k45 = 0.565

Table 7.3 Pspice inductance values of tracker power
cable - MSC (LV and HV wires).
Capacitance

nf
----m

C11

0.163

6.52

nF

9.78

nF

C22

0.187

7.48

nF

11.22

nF

C33

0.291

11.64

nF

17.46

nF

C44

0.061

2.44

nF

3.66

nF

C55

0.062

2.48

nF

3.72

nF

C12

0.040

1.6

nF

2.4

nF

C13

0.046

1.84

nF

2.76

nF

C14

0

0

nF

0

nF

C15

0

0

nF

0

nF

C23

0.005

0.22

nF

0.33

nF

C24

0

0

nF

0

nF

C25

0

0

nF

0

nF

C34

0

0

nF

0

nF

C35

0

0

nF

0

nF

C45

0.0183

0.732

1.09

nF

Section 1 - 40 m Pspice
lump values

nF

Section 2 - 60 m Pspice
lump values

Table 7.4 Pspice capacitance values of tracker power
cable - MSC (LV and HV wires).

249

7. Susceptibility to transient perturbation

ResistancemΩ--------m

R11

3.15

0.126 / 2Ω

0.95

0.067 / 2

Ω

R22

4.65

0.186 / 2Ω

1.78

0.125 / 2

Ω

R33

1.75

0.070 / 2Ω

0.68

0.044 / 2

Ω

R44

24

0.96 / 2Ω

24

1.44 / 2

Ω

R55

26

1.04 / 2Ω26

1.44 / 2Ω

Section 1 - 40 m
Pspice lump valuesmΩ--------m

Section 2 - 60 m Pspice
lump value

Table 7.5 Pspice resistance values of tracker power
cable - MSC (LV and HV wires).
Figure 7.6 shows the Pspice model used in the tracker transient analysis.

Figure 7.6: Tracker power supply distribution - Pspice circuit.

250

7.3. Surge or slow transients

7.3.2.1.1 Sudden load changes
The first analysis presented corresponds to a sudden load change in the LV line. The
power consumption in the V2.5 lines drops from 11 A to 9 A, and from 5 A to 0 A in
the V1.25 line. This change of load produces an over voltage at the end of the line
(FEE), generated by the energy stored in the cable and the high value of the voltageof the power supplies, which is necessary to compensate the voltage drop along thecables. The reaction of the closed-loop system against this transient cannot be fasterthan hundreds of µ s because of the long length of the cable and the high value ofthe filter capacitors limits the bandwidth of the system. Figure 7.7 shows thetransient generated in the LV lines under such conditions. The voltage at the FEElevel pass the maximum over voltage recommended by the manufacturer of the
APV25, which is around 2.8 V, producing the destruction of the FEE.
5.0V

4.0V

3.0V

2.0V

1.0V

0V
0s
V(I2:+)

100us
V(I1:+)

200us

300us

400us

500us

600us

700us

Time

Figure 7.7: Simulation results of Over voltage on the V1.25 and V2.5 line in the Tracker
sub-system generated by a load change of 11A to 9A-V2.5 line and 5 to 0 -V1.25 line.
One possible solution to avoid the destruction of the FEE by over voltage could be
to open the power supplies when a transient is detected, and leave the circuit inopen loop (un-regulated mode). However, the high value of the voltage of thepower supplies necessary to compensate the voltage drop along the LV cables forces
the transient to pass above the maximum value valid for the FEE. This phenomenon
appears also in the case that bigger values of the capacitor at the input of the FEE
are chosen. Figure 7.8 shows the over voltage generated for different values of the
FEE capacitors (Cb = 483 µ f , Cb = 600 µ f and Cb = 1000 µ f ).

251

7. Susceptibility to transient perturbation

4.0V

3.0V

2.0V

1.0V

0V
0s

100us
V(I2:+)

200us

300us

400us

500us

600us

700us

V(I1:+)
Time

Figure 7.8: Simulation results of over voltage on the V1.25 and V2.5 line in the tracker
sub-system generated by a load change of 11A to 9A-V2.5 line and 5 to 0 -V1.25 line fordifferent values of FEE capacitors.
Other possible solution could be to place TVS (transient voltage suppressors) toabsorb the transient and avoid the damage of the electronics. The energy that thisdevice has to absorb during the transient is relatively small. However, there are twovery important constraints that make it quite difficult to use them. First of all, the
TVS should have a nominal voltage of 2.5 V, with maximum clamp voltage of 2.8 V,
and second, this device has to be a RAD tolerant device, as the tracker is placed in
an area with high radiation. Both characteristics force the device to be a custom
design.
The problems pointed above force us to find other solutions to suppress thetransients. The solution has to be focused on the design of power supply unit. One
possible solution could be to use 2 quadrant power supply units. This type of units
allows to apply a negative voltage at the output of the power supply when atransient is detected after a sudden change of load. This negative voltage accelerates
the discharge of all energy stored in the cable decreasing, substantially, themaximum value of the voltage transient. Another solution, focused on the design ofthe power supply unit, could be to change the capacitor at the output of the power
supply to a very small one (lower than 1 µ F ). The function of this capacitor is tochange the resonance frequency of the circuit. When a change of load is detected,
the power supply unit opens and the cable inductance resonates with this capacitor.
If a large value is selected for this capacitor (as the previous example presented), theover voltage pass the maximum value valid for the FEE. However, if a small value isselected for this capacitor, the transient response of the cable is fast discharging theexcess of energy and keeping the maximum value of the transient voltage at the
input of the FEE lower than the maximum value tolerated. Figure 7.9 shows the252

7.3. Surge or slow transients

evolution of the voltage on the FEE for different values of the output capacitor of
the power supply unit. (Cps = 1 µ f , Cps = 100 µ f and Cps = 500 µ f ). This solution
can be used in combination with a resistance at the power supply, which discharges
the remaining energy in the system during the last section of the transient.
4.0V

3.0V

2.0V

1.0V

0V
0s

100us
V(I2:+)

200us

300us

400us

500us

600us

700us

V(I1:+)
Time

Figure 7.9: Simulation results of over voltage on the V1.25 and V2.5 line in the tracker
sub-system generated by a load change of 11A to 9A-V2.5 line and 5 to 0 -V1.25 line fordifferent values of PS capacitors.
7.3.2.1.2 Magnetic coupling transients
Cables or bundles of cables, which are placed in the same area, are susceptible toover-voltages induced by high current changes in one or more of this cables. These
currents produce low frequency magnetic fields, which can be coupled to adjacentcables causing bad performance of the equipment connected to these cables.
Different short-circuits in the HV system of the tracker sub-detector has beenchosen to present the problems induced by these transients. When a short circuit
occurs in the HV system, the PS unit reacts very fast and opens the power units.
From that moment, the HV circuit becomes an un-regulated system in which thepower supply is represented by the output capacitor, which is charged to 600 V. This
circuit has been implemented in Pspice (figure 7.6) to study the influence of two
different HV short circuits in the distribution line.
The first case corresponds to a balanced short circuit between both HV cables at the
input of the HV-FEE (a short circuit in the HV input capacitor). In this case the shortcircuit current returns to the power supply through the HV return line of the cable.
The second transient corresponds to an unbalanced short circuit at the input of the
HV-FEE. In this transient the current cannot return to the power supply through thesame cable1. This kind of transient is not frequent and depends on the granularityand grounding layout of the tracker detector, which has not been decided yet.

253

7. Susceptibility to transient perturbation

However, for the present example it has been considered as a potential problem
because it helps to understand better the problem of mixing cables, carrying
different current levels in the same bundle.
The simulated model corresponds to a PS distribution unit of layer 4 of the tracker
outer barrel (TOB). This unit supplies power to 24 APVs and the nominal current
consumption per line is approximately 5 A for V2.5 line and 1 A for V1.25 line. The
FEE-hybrid LV channel is simplified by two identical input module capacitors(Cb=160 µ f ) and by a current source in parallel with a damping resistance. The
damping resistance for the model in the V2.5 line is equal two 1.1 Ω and for theV1.25 is equal to 1 M Ω . Figure 7.10 and 7.11 represent the short circuit current in
the HV line and the induced voltage at the 2.5V and 1.25V lines for balanced andunbalanced short-circuits, respectively.
40A
30A
20A
10A
0A
-10 A
-20 A
A V G X (-I( L 4 A ) ,5 u s )3 .0 V

2 .0 V

1 .0 V

SEL>>
0V
0s

50us
V ( I1 :+ )

100us

150us

200us

250us

V (I 2 :+ )T im eFigure 7.10: Balanced short circuit current in the HV line and voltage at the 2.5V and
1.25V lines at the input of the FEE
30A
20A
10A
0A
-1 0 ASEL>>-2 0 AA V G X ( -I ( L 4 A ) ,5 u s )4 .0 V

3 .0 V

2 .0 V

1 .0 V0V
0s

50us
A V G X ( V ( I 2 :+ ) , 5 u s )100us
A V G X (V ( I 1 :+ ) ,5 u s )

150us

200us

250us

300us

T im eFigure 7.11: Unbalanced short circuit current in the HV line and voltage at the 2.5V and
1.25V lines at the input f the FEE.
1 In this case the current ’return’ presents the same resistance as the normal one tohave the possibility to compare the effects in both configurations.
254

7.3. Surge or slow transients

This transient cannot be easily cancelled using TVS or regulated by the PS units, as
it was the case in the previous section. The easiest way to control the magnitude of
the transient is to limit the coupling energy involved in the transient, which isdefined by the capacitor at the output of the HV power supply unit and thegeometrical configuration of the wires that belong to the same bundle. The
geometry of the cables sets the value of the mutual inductance and defines theamount of energy coupled from the primary circuit (noise source - HV circuit) to thesecondary circuit (noise victim - LV circuit). It is always recommended to separate
cable bundles that work with different level of power.
The capacitor at the output of the power supply defines the maximum amount ofenergy present in the transient. This energy corresponds to the energy stored in thecapacitor as it is shown in equation 7.4. In the present case, the HV is constant andequals to 600 V. The value of the HV output capacitor limits the value of this energy.
21
Ec = --- ⋅ C ⋅ V
2

(7.4)

Figure 7.12 shows the maximum voltage in the V2.5 line and maximum HV shortcircuit current for different values of the HV output capacitor for a balanced (figure
7.12a) and unbalanced short-circuit (7.12b).

a

b

50A

50A

40A

40A

30A

30A

20A

20A

10A

10A

0A
0.1u
0.2u
0.3u
Max(AVGX(-I(L4A),5us))

0.4u

0.5u

0.6u

0.7u

0.8u

0.9u

1.0u

Cc

0A
0.1u
0.2u
0.3u
Max(AVGX(-I(L4A),5us))
5.0V

4.0V

4.0V

3.0V

3.0V

0.4u

0.5u

0.6u
Cc

0.5u

0.6u

0.7u

0.8u

0.9u

1.0u

0.6u

0.7u

0.8u

0.9u

1.0u

Cc

5.0V

SEL>>
2.0V
0.1u
0.2u
0.3u
Max(AVGX(V(I1:+),5us))

0.4u

0.7u

0.8u

0.9u

1.0u

SEL>>
2.0V
0.1u
0.2u
0.3u
Max(AVGX(V(I1:+),5us))

0.4u

0.5u
Cc

Figure 7.12: Maximum voltage on the 2.5 V line and shortcircuit current in the HV line fordifferent values of HV output capacitor (balanced and unbalanced shortcircuit).
The worst case corresponds to the un-balanced short circuit. For the present
example, limiting the value of the capacitor could be good enough to protect oursystem. However in some cases if the neighbor circuit presents high-energy
transients (power line with fuses), the energy involved in the transient may behigher than the previous case, resulting in a serious problem for the detector. This255

7. Susceptibility to transient perturbation

issue can be solved grouping the cables in bundles according to the power leveltransmitted and separating these bundles by a safe distance to ensure a low energy
coupling among cables. In the literature it is possible to find a variety of papers[12][13][14] where this problem is studied in detail.
7.3.2.2 Multi-module DC distribution
Transients generated in a generic DC low-voltage distribution system are studied inthis section. The topology analyzed corresponds to a distribution line carrying highcurrent to a distribution bus where several groups of FEE are connected to the bus
in parallel. This study addresses the implications of the detector granularity in thedesign of power supply distribution system and the selection of protection devices.
7.3.2.2.1 Model
As example of a generic power supply distribution system, we choose one whichfeeds 4 groups of generic FEE units at + 6.5 V. The power is supplied to the FEE unit
through a long shielded power cable of 20 meters. Each FEE group consumes thesame amount of current. For the present study, we assume that a short-circuit
occurs at the input of one of the units. This failure blows the fuse protecting thefaulted path. The length of the line is varied between 20 m, 50 m and 100 m, and thepower consumption per unit changes between 5 A, 15 A and 25 A (total current - 20
A / 60 A / 100 A) in order to study the influence of these parameters in the shortcircuit characteristics (energy, short circuit current and duration). A schematic of themodel presented is shown in figure 7.13.

TVS
Short !!!

Input Capacitor

Figure 7.13: Generic power supply distribution system - Pspice circuit.

2567.3. Surge or slow transients

Although there are some papers [15] defining some fuse models for over current
protection, we simulate during this study the fuse operation by means of switching
devices included in the Pspice libraries. A close-switch is used to create the shortcircuit and a open-switch is used to simulate the fuse opening. A delay between
both switches is included to simulate the opening time of the fuse. The opening
time and resistance of the fuses are selected based on real fuse characteristics. The
fuses selected (257075, 257020 and 257040) are fast acting fuses types from Littlefuse
Company and the main characteristics of these fuses are given in appendix B.
Each FEE group is simplified by a current source in parallel by a damping resistanceand a capacitor. The damping resistance for the model is equal to 80 Ω and thecapacitor is equal to 30 µ F . At the distribution bus a filter with several capacitors of3000 µ F is placed to remove part of the perturbing signals on the power supply
line. The number of these capacitors varies between 1 for the 20 A line, 2 for the 60
A line and 4 for the 100 A line. The parasitic components of this capacitor have been
included in the model. Their values are based on measurements performed on realfilter capacitors of the HCAL sub-system.
The duration of theses transients are of the order of µ s , which allows to use thelumped model for the distribution line presented in the previous section. The cablemodel used corresponds to the HCAL power cable. The per-unit length parametersof the cable are calculated based on measurements performed at 10 kHz and used in
the Pspice simulation.
During the present study the per unit length resistance values of the cable aremodified when the length of the cable and/or the current increases in order to keepa constant voltage drop of 1.4 volts along the cables. That means that the section ofcable is changed, accordingly, but the per unit length values of L and C keeps
constant. This assumption is not true in real applications because if the section ofcable changes the geometric configuration of cable changes and hence the per unit
length values of the L and C. Nevertheless this assumption can be considered valid
due to the goal of the study is to show the problematic with different configurations
and not to estimate the exact value of the transient. The per unit length L, R, C and
Pspice parameters are given in tables 7.6 to 7.7 and 7.8.
InductanceµH-------m

20m

50m

100m

L11

0.24334.686------------- µ H
211.715---------------- µ H2

23.43------------- µ H
2

L220.24264.852------------- µ H
2

12.13------------- µ H
2

24.26------------- µ H
2

L120.0097k12=0.04

k12=0.04

k12=0.04

Table 7.6 Pspice and per-unit length inductance valuesof the power distribution cable.

257

7. Susceptibility to transient perturbation

CapacitancenF------m

C11

0.1880

3.76

9.4

18.8

C22

0.1927

3.854

9.635

19.27

C12

0.0412

0.824

2.06

4.12

20 m

nF

50 m

nF

100 m

nF

Table 7.7 Per unit length and Pspice capacitance valuesof the power distribution cable.

Resistance( Ω /km)

20 m

50 m

100 m

20 A

1.75

0.7

0.35

60 A

0.583

0.233

0.116

100 A

0.35

0.14

0.07

Table 7.8 Per unit length resistance values of the power
distribution cable.

Resistance(R11 = R22)

All lengths

20 A

0.035
------------- Ω
2

60 A

0.011
------------- Ω
2

100 A

0.007
------------- Ω2Table 7.9 Pspice resistance lumped values of power
distribution cable.
7.3.2.2.2 Short circuit transient
When a short circuit occurs in one FEE group, major disturbances appear on thecable and at the distribution bus bar level. The current in the cable increases up to a
maximum value fixed by the resistance of the cable and the short circuit contact
resistance. Afterwards, for a short period of time this current remains steady in the258

7.3. Surge or slow transients

cable until the fuse melts. This time depends on the clearing time of the fuse, which
depends on the fuse selected. During this period, the high current causes a voltage
dip at the input distribution board, where all other FEE groups are connected. Later,when the fuse melts, the cable current decreases and an over voltage appears at the
input distribution board. This over voltage may cause malfunction or thedestruction of the FEE connected to the same input distribution bus. Figure 7.14
shows the current in the cable and the distribution bus voltage during the shortcircuit for the case of a cable of 20 meters, carrying a nominal current of 60 amps.

300A

200A

Short circuit starts

100A

The fuse melts
0A
I(L1B)
12V

Over-voltage
8V

Voltage dip

4V

SEL>>
0V
0s

2ms

4ms

6ms

8ms

10ms

12ms

14ms

V(U22:1)
Time

16ms

18ms

20ms

Final stage

Figure 7.14: Cable current and FEE input voltage during a short circuit.
Several aspects are considered to characterize the transient during a short circuit:• Short circuit current and voltage dip value.
• Over voltage (Value & duration).
• Excess of stored energy involved when the fuse melts.
The over voltage waveform at the time the fuse opens is characterized by its
maximum value and duration. This duration is defined by the rise time value (tr)and half time value (t50%), which is considered as the time difference between the
moment in which the fuse opens and the moment in which the voltage decays to
50% of the voltage difference between maximum voltage and the final voltage.
These values are given in detail figure 7.15.

259

7. Susceptibility to transient perturbation

11.50V

Vmax

10.00V

50% - (Vmax-Vfinal)
8.00V

90% - (Vmax-Vdip)
6.00V

Vfinal
4.00V

10% - (Vmax-Vdip)

Vdip
2.00V

0.76V
16.000ms
V(U22:1)

tr

16.200ms

16.400ms

t50%

16.600ms

16.800ms

16.932ms

Time

Figure 7.15: Over-voltage parameters
The excess of energy involved during the transient when the fuse opens helps todefine and select the transient voltage suppressor device (TVS). This energy isdefined as the difference between the energy stored in the cable and in the
distribution bus capacitors when the fuse opens and the final stored energy insteady state.

1
1 2
1
1 2
Exc -E = ⎛ --- ⋅ I cc ⋅ L + --- ⋅ V dip ⋅ Cd⎞ – ⎛ --- ⋅ I f ⋅ L + --- ⋅ V f ⋅ C d⎞
⎝2
⎠ ⎝2
⎠
2
2

(7.5)

where;•

I cc is the short circuit current.

• V dip is the voltage dip at the distribution bus.
•

I f is the current in the cable during the final steady state.

•

V f is the distribution bus voltage during the final steady state.

•

Cd is the total capacitance of the distribution board• L is the cable inductance, which is defined in this circuit as:

L = ( L 11 + L 22 – ( 2 ⋅ L 12 ) ) ⋅ Lg

(7.6)

where Lg is the cable length. Table 7.10 summarizes all the parameters mentioned
before:260

7.3. Surge or slow transients

Cable
Current
Cable
length

20m

50m

100m

Distribution board voltage

Energy

Io
(A)

Icc
(A)

If
(A)

Umax
(V)

tr
µs

t50%
µs

Vdip
(V)

Vf
(V)

Exc.E
(mJ)

20

102

15

7.3

13

47

0.75

6.84

-

60

293

45

11.25

9.27

322

1.06

6.84

245

100

494

75

11.33

59

724

0.98

6.84

813

20

102

15

8.07

18

564

0.75

6.84

46

60

293

45

14.35

107

807

1.06

6.84

817

100

494

75

16.84

295

1240

0.98

6.84

2438

20

102

15

10.49

100

903

0.75

6.84

163

60

293

45

19.52

289

1220

1.06

6.84

1776

100

494

75

23.7

579

1810

0.98

6.84

5121

Table 7.10 Transient parameters during a short circuitin the generic PS distribution system
Based on the values shown in the table, it can be observed that the layout andgranularity of the detector has a big impact in the transients generated by fuses inthe detector. For a constant length, the higher the nominal current, the higher are
over voltage and stored energy during the fault. For a constant current, the longer
the length, the higher are the over voltage and stored energy during the fault.
The protection of the system against the over-voltage transients can be achieved by
TVS for some sub-systems located in areas where the radiation level is not very highand the over-voltage margin of the FEE is relative large. The TVS is a siliconavalanche device and the more important parameters in its selection are:
• Reverse standoff voltage (Vr). This parameter should be equal to, or larger than,the peak operating level of the circuit (or part of the circuit) to be protected.
• The peak pulse current (Ip) is the maximum current that the TVS can withstandwithout damage. This value can be determined by dividing the peak transient
voltage by the source impedance.
• The maximum clamping voltage (Vc) is the peak voltage that will appear acrossthe TVS when subjected to the peak pulse current (Ip), based on a 1 ms
exponential. This waveform is a 10/1000 microsecond waveform. Thisparameter basically fixes the maximum energy absorbed by the TVS.

261

7. Susceptibility to transient perturbation

The selection of the TVS involves the knowledge of the parameters such as limiting
voltage, surge current and maximum energy dissipation.
The TVS data sheet, generally, are referred to a specific surge test that should betaken into account to select the TVS that fits better to our requirements. Based on
the result shown in table 7.10 two TVS of 1500W and 5000 W with a reverse
stand-off voltage of 6.5V (1.5KE7.5A and 5KP6.5A) from Littlefuse are selected forthis analysis. The main characteristics of both TVS are for the 1500W device a
maximum surge current of 132A and maximum clamping voltage of 11.3V referred
to a current waveform of 10/1000 µ s , while for the 5000W device a maximum
surge current of 407A and maximum clamping voltage of 12.3 V referred to thesame current waveform. These waveforms describe maximum abortion energy ofthese devices of 2 Jules and 7 Joules, respectively, which fits quite well with therequirements of the transients presented in the simulation model.
The circuit is analyzed including the selected TVS to verify the validity of theselection criteria presented above. For that purpose the TVS is modeled in Pspicecircuit including its parasitic components. It can be modeled [16] using a resistanceand inductance in series with a battery in parallel with a capacitor. The simulations
presented have been performed with the first TVS selected, the 1.5KE7.5A. The
model parameters of this TVS are R ta = 1.9 m Ω , L ta = 11 nH , C ta = 10 pf andV ta = 6.8 V . The Pspice model validation [17][18] is given in appendix B. During
this simulation, the current through the TVS, the voltage, the power and the energy
and the voltage have been calculated using Pspice simulation.
Figure 7.16 shows the simulated current and the voltage supported by the TVS
during a short-circuit in the system in which the length of the cable is 50m and thenominal current is 60 A.
150 V

100 V

50V

0V
AVG X(I(Vta),5u s)
10V
8V
6V
4V
2V
SEL >>
0V
0s

5 ms

10ms

15ms

20ms

25m s

AVG X(V(C2a1c:2),5 us)
T ime

Figure 7.16: Current and voltage in the TVS during a short circuit for a 50 m long cableand nominal current of 60 A.

262

7.3. Surge or slow transients

Table 7.11 summarizes the simulated data of the TVS-1.5KE7.5A performance forsome of fault cases analyzed previously.
Cable current
Cable
length

20m

50m
100m

TVS performance data

Io(A)

Icc (A)

Imax
(A)

Vcmax
(V)

Pmax
(W)

Energydissipated
TVS (mJ)

60

293

80

8.13

610

205

100

494

118

8.42

1000

712

20

102

6

7.57

45

16.5

60

293

112

8.38

945

913

20

102

20

7.76

194

155

Table 7.11 TVS performance data during the shortcircuit.
The presence of high currents during the short circuit may be solved using current
limiters or electronic fuses, which limits the maximum value of the current during
the short circuit to a 30-50% of the nominal current. Table 7.12 summarizes theenergy involved during the transient if fuses are replaced by current limitators.

Cable Current
Cable length

20m

50m

100m

EnergyIo (A)Icc (A)

Exc.E (mJ)

20

26

-

60

78

-

100

130

-

20

26

-

60

78

16.24

100

130

69.41

20

26

-

60

78

63.19

100

130

198.44

Table 7.12 Transient parameters during the shortcircuit in the generic PS distribution system withcurrent limitators.

263

7. Susceptibility to transient perturbation

Based on the previous analysis, some power distribution layouts are characterized
by large currents and stored energy during the short-circuit. The use of TVS as the
only solution to avoid over-voltages on the power distribution lines is not probably
the best strategy considering the radiation environment and the reduced space tolocate the protection device. In those cases, the selection of an optimal granularity ofthe detector in combination with electronic fuses can avoid the use of large TVS.

7.3.3 Surge test for the CMS detector.
7.3.3.1 Introduction
Surge immunity tests are used to evaluate the performance of the equipment inpresence of slow transients applied to the input/output cables and to define theprotection devices necessary to attenuate its effects into the equipment. The surge
immunity tests use generators with defined waveforms and test set-ups to achieve
repeatable tests [19][20][21] that are representative of a set of characteristic transientthat appears in the system. It is important to realize that these tests do notnecessarily simulate exactly the real electromagnetic disturbances present in thesystem but they give the necessary information about its effect in the equipment toqualify its performance and robustness to a set of transient perturbations.
There are a great variety of standards that defines different normalized proceduresto evaluate the immunity of the equipment to these transients. The two mostcommon tests are defined by the standards EN-61000-4-5 and ANSI/IEEE
C62.41-1980 [22]. These standard procedures define three normalized waveforms,
which are divided in bidirectional waveforms and unidirectional waveforms.
Figure 7.17 shows the wave-shape of a bidirectional waveform. This wave is basedon the original voltage wave that was proposed by ANSI/IEEE C62.41-1980. It isrepresentative of general surge voltages appearing in indoor power distribution
systems. This wave shape rises in 0.5 µ s and decays while oscillating at 100 kHz(’called ring wave’), each peak being about 60% of the presiding peak.

Figure 7.17: Bidirectional waveform for the surge immunity test.

2647.3. Surge or slow transients

The unidirectional waveform is more typical for outdoor or unprotected systems,
where the attenuation has not taken place and substantial energy or current is stillavailable. One of these unidirectional impulse waveforms specified by thestandards is shown in figure 7.18. This impulse is defined by the combination of
two waveforms, 1.2/50 µ s voltage impulse in open-circuit and a 8/20 µ s current
impulse into a short circuit, leading to its common name: ’the combination wave’.

Figure 7.18: Unidirectional waveform for the surge immunity test.
The other unidirectional waveform is similar to the previous one but this is only
specified by the open circuit voltage (10/700 µ s ) according to the standard
publications [9].
A surge generator is used to inject the transient to the circuit. A
coupling-decoupling network (CDN), which is located in the generator, is used totransfer the energy into the circuit. The amount of energy and waveformcharacteristic depends on the elements of this network as well as the internal
impedance of the generator. Therefore all the components of the CDN and theinternal impedance of the device are specified by the standards [9]. The CDN has anadditional function that is protecting the auxiliary equipment used in the test andensuring the pulse go into the equipment under test and not to other parts of thesystem Figure 7.19 shows the set-up used in this test.

Figure 7.19: Typical set-up for the surge immunity test

265

7. Susceptibility to transient perturbation
Typical values of these network components are generally a capacitor of 9 µ F or 18µ F for the capacitor, 10 Ω or 40 Ω for the resistor and an inductance of somemH.
The normalized waveform to be applied during the tests of the sub-detector
electronics has to be representative of the transient perturbations characteristic ofthe different areas of the CMS detector. The definition of this normalized waveform
to be used is important for the design of protective devices and also will define thereliability and cost of the system. Wrong surge test characteristics, like excessive
duration of the perturbation, large voltage perturbation or very low source
impedance, force a high-energy requirement on the protection device or transient
suppressor.
7.3.3.2 Characterization of the HCAL FEE to surge test.
The goal of this test is to study the effect on the internal voltage distribution of surge
perturbations. The FEE used for RF conducted noise immunity test has not beenconnected to the backplane because, at the time the test was performed, thedestruction of this equipment implied serious problems for the project as it was the
only prototype of the HCAL FEE.

Surge
Generator

I1 Input filter

TVS-1500W
Cf=3000µf

CDN
5.5V

V2
UCS 500

6.5V

3.5V
LinearRegulator

V1

Backplane

Rload

Cb=240µf

Surge Test Layout

Dummy board

Figure 7.20: Surge test layout for the HCAL FEE.
Figure 7.20 shows the test layout. Unidirectional 1.2/50 voltage impulse waveformsof 200V peak with both positive and negative polarities in DM and CM
configuration is applied to the system where the FEE is now replaced by dummyboards. Each dummy board consists of filter capacitors, the LV regulator andresistors to simulate the consume of the FEE board. The input power filter has a266

7.3. Surge or slow transientstransient voltage suppressor (TVS) of 1500 W and nominal voltage of 7.5 V. The
backplane voltage (V1), the voltage at the output of the LV regulator (V2) as well as
the current injected into the system (I1) has been measured using a digital Tektronix
scope and a differential voltage probe. Figure 7.21 depicts the dummy board, while
figure 7.22 shows a picture of the RBX layout during the test.

Connectors
LV regulator

Resistance

Figure 7.21: Dummy board layout during HCAL-FEE surge test.

Backplane

Input filter

RBX

Connectors

Copper plane
Figure 7.22: RBX layout and input filter during HCAL-FEE surge test.
7.3.3.2.1 DM surge immunity test.
A voltage surge of 200 V with both positive and negative polarities is applied to thesystem. The voltages are applied in DM and the current is injected through acapacitor of 18 µ f without any resistance in series. This makes the total value of the267

7. Susceptibility to transient perturbation

source impedance equal to 2 ohms. The current injected has a peak of 85 A for the
positive polarity and 83 A for the negative one, as it is shown in figure 7.23.

Figure 7.23: Current injected in the HCAL-FEE prototype during the DM surge test
(positive and negative polarities).
The voltages at the input of the dummy board (back-plane voltage) and the outputof the LV regulator when the transient is applied are shown in figure 7.24. For the
positive voltage surge the system is protected without any problem. The TVS andthe HF capacitor placed on the back-plane limit the peak voltage in the back-plane
to 8 volts. The transient voltage is completely attenuated by the LV linear regulator
giving clean 3.5V to bias the digital electronics.

Figure 7.24: Backplane and regulator output voltage during the positive DM surge test.
For a surge with negative polarity (figure 7.25) the TVS and the voltage regulator do
not work, generating on the 3.5V line a voltage dip of short duration, which could
produce malfunction in the digital section.

268

7.3. Surge or slow transients

Figure 7.25: Backplane and regulator output voltage during the negative DM surge test
These effects may constitute a serious problem in the HCAL-FEE, however it is notclear that these transients appear in the CMS detector. Also, it is important tomention, that there are some TVS, which work with both polarities and the use of
these devices may solve the problem.
7.3.3.2.2 CM surge immunity test.
A voltage surge of 200 V with both positive and negative polarities is applied to thesystem. The voltage is injected through a capacitor of 9 µ f with a resistance inseries of 10 ohms. This makes the total value of the source impedance equal to 12
ohms. The current injected has a peak of 15 A for the positive polarization, as it isshown in figure 7.26.

Figure 7.26: Current injected in the HCAL-FEE prototype during the CM surge test
(positive polarity).

269

7. Susceptibility to transient perturbation

It is observed that for the same voltage applied the higher the value of the
impedance the lower the value of the current. The current limitation generated by
the impedance is important as it limits the over voltage generated by the current
injected. This limitation causes a very low over voltage, which does not force the
TVS to work and the transient is easily absorbed by the LV regulator. Figure 7.27
shows the backplane and linear regulator output voltage when a surge of 200 V in
CM is applied to the dummy board.

Figure 7.27: Backplane and regulator output voltage during the positive CM surge test
7.3.3.3 Protection stages for the CMS detector.
Surge transients can easily occur in the CMS detector. The amplitude, polarity andduration depend on the configuration and the layout of the power supply
distribution system. It has an important impact in the selection of the protections of
the system since TVSs are not ideal devices and present some parasitic parameters,
which limit the amplitude of the current, power and energy absorbed by the TVS. In
addition, as it has been presented in the previous sections, the transient is not totallyabsorbed by the TVS. The ceramic capacitors on the FEE and the LV linear
regulators help to deal with these transients. This introduces an important point in
the design of protections against transients, which is the protection stage.
If a transient cannot be attenuated using a single component, it is possible tocombine different components and utilize their respective advantages to limit thetransient in several steps. Figure 7.28 shows an example of the principle of staged
protections. This example helps to limit an over voltage of 10 kV in three stages.

270

7.4. Burst or fast transients

Figure 7.28: Protection stages against a surge of 10 kV
A similar system can be used to protect the FEE of the CMS detector. The firstprotection stage can be made at the counting room level, where TVS (surge arrestors
or varistors), which are able to deal with high energy transients, can be placed. This
device protects the AC-DC converters and DC-DC converters from the transients
generated in the AC-400Hz line. In the case to distribute directly AC power to theperiphery of the detector, these devices have to be placed on the periphery of thedetector. A second protection stage is focused to protect directly the FEE. A
combination of TVS, ceramic capacitors and LV linear regulators form this second
protection unit. It focuses on the protection of the DC power supply line. However abad selection on the distribution layout (cables bundle distributions and power
supplies location), granularity (lines with high values of nominal current) andshort-circuit protections (no current limitators), make it almost impossible to protectof the FEE against transients presented on this second part of the system. This will
have a very important impact in terms of reliability and cost of the system.

7.4 Burst or fast transients
Electric arcs are generated between the mechanical contacts of circuit-breakers,
when inductive loads like relay coils, timers, motors contactors etc. are connected or
disconnected from the line. The over voltage generated between the contacts during
this transient can be higher than the isolation voltage thus producing the spark
re-ignition. Arc ignition and interruption leads to a burst process. This intermittent
process finishes when the voltage stress presented on the switching contacts is
below the electrical and thermal strength of the spark gap channel. Figure 7.29
shows the fast transient switching process.

271

7. Susceptibility to transient perturbation

Figure 7.29: Burst effect.
This effect produces a repetitive and fast over voltage that propagates through thesystem and can affect the electronics devices connected to this unit. This transient is
not especially important for its destructive effects but for the HF spectra content of
the signal that can produce disturbances in safety devices (as electronic fuses,
measurement devices or alarms) with its posterior consequences.
The fast transients contain high frequency harmonic components, which are
expected to be easily attenuated as they propagate through the cables. In the CMS
detector, the final location of electro-mechanical switches is not yet defined. A
possible location is in the counting room (120 meters away from the detector) or at
surface level (around 250 m far from the detector). Based on some studies presented
in [23] these distances can produce between a 50% and 80% of attenuation of theamplitude of fast transients. This reduction in combination with the power level ofthese transients will produce very low and slow burst transients that can be easily
filtered by HF capacitors and the LV regulator at the FEE board.
The EMC standards EN-6000-4 [24] define some test procedures to ensure the good
performance of the system against this type of transients. The standard waveform
into the 50 ohms load condition is characterized by a very fast rise time (5 ns )followed by a decay time of 50 ns . The test levels varies from 250 V to 5 kV andburst duration and period are also fixed by them, having values of 15 ms and 300ms
respectively. Finally the repetition rate of the pulse is also defined by the standards
and varies between 5 kHz and 2.5 kHz. The energy is injected into the systemthrough a capacitor of 33 nF and the maximum energy injected into the system islimited to 4 mJ/pulse. Figure 7.30 shows the waveform defined by European
standards.

272

7.5. Quenching effects

Figure 7.30: Burst test standard waveform.

7.5 Quenching effects
The tracker, ECAL and HCAL electronic systems are placed inside thesuperconducting solenoid1, which forms a uniform magnetic field of 4 Tesla. Thissolenoid of 6 meters of diameter and 13 meters of length, has a nominal current of19500 A and stores around 2.7 GJ of energy during normal operation. The
inductance of the solenoid is 14.15 H
The charge and discharge of the magnet under normal conditions requires several
hours and should not present any problem for the electronics devices. In two special
cases the discharge of the magnet is performed in a relatively short period (10minutes). These are the fast discharge and the quench discharge. The analysis of theeffects induced in the CMS detector by these transients and the associated induced
voltages by the magnet quenching are presented in this section.
A quench is known as the transition from the superconducting state to the normal
conducting state of the conductor. If this phenomenon happens in the
superconducting solenoid it is necessary to dissipate as fast as possible all theenergy stored in the magnet to avoid the destruction of the superconducting cable.
Figure 7.31 shows the discharging time of the magnet during a quench [25].

1 Chapter 1 - Section 1.6

273

7. Susceptibility to transient perturbation.

Figure 7.31: Solenoid current during an accidental quench.
To estimate the voltage induced in a certain region during a quench, the magnetic
field value at an internal point of the solenoid, far away from the ends, induced by
the solenoid current can be simplified as:

N ⋅ i ( t ) Wb
B ( t ) = µo ⋅ ----------------- -------2Lg
m

(7.7)

where N is number of turns of the solenoid, Lg the length of the solenoid, therelative permeability of the media is 2 µ r = 2 and i(t) the current flowing throughthe solenoid. This field is calculated as:

N ⋅ i ( t ) Wb
–72168
B Th = µ o ⋅ µ r ⋅ ----------------- -------2- = 4 ⋅ π ×10 ⋅ 2 ⋅ ------------ ≈ 4 Tesla
Lg m
12.5

(7.8)

Assuming that in a specific area (S) perpendicular to the magnetic field the field isconstant and the inducted voltage is:Φ = ∫ B ds = B ⋅ S

(7.9)

dΦ
dB
= –S ⋅
dt
dt(7.10)

V = –Combining both equations lead to voltage per unit area

N d
V = K ⋅ µ ⋅ ------o Lg- ⋅ d t i ( t )S(7.11)

Figure 7.32 shows the voltage induced per square meter in different areas of thedetector (central part and at the end of the solenoid). The maximum voltage274

7.6. Summary

induced corresponds to the central part of the detector which is the 4 tesla area andmVthis value is around 43 -------2- . For the end of the solenoid this value is around 21.5m
mV-------2- .
mFigure 7.32: Voltage induced per squared meter in the internal and end part of the solenoid
These values are very low and may not cause any problem to the electronics
circuits. However, the detector has several short-circuited structures, like pipes,
copper or aluminum blocks, where part of the electronics are locally grounded(signal reference ground). Large currents may be induced in these blocks that may
result in a damage of the electronics connected to these blocks.

7.6 Summary
Transients in power distribution lines can be generated by variations of the current
consumption of the FEE due to load changes, random failures, turn On/Off ofequipment, etc. They induce over-voltages and voltage dips in the LV-DC and the
AC distribution system, which can affect the performance and the reliability of the
FEE. The FEE immunity analysis to voltage dips, surge transients and fast transientshas been analyzed in the present chapter to define the base line of a future EMC
policy for the detector.
Voltage dip tests were conducted applying short interruptions and voltage
variations to the HCAL-FEE and evaluating the degradation of the FEE
performance and the loss of acquired data. These tests have shown that the FEEshows a gain loss of the amplifier when the perturbation is applied to the analogue
power line, while when voltage dips are applied to the digital power line, thedigital part of the FEE is very sensitive to these transients forcing to reset the system
when the perturbation reaches the limit values.

275

7. Susceptibility to transient perturbation

The surge transient studies and tests have been focused on the characterization of
the protection devices to be included in the FEE. The CMS detector can be divided
in two areas; one corresponding to the AC power supply distribution system andthe other corresponding to the DC power supply distribution system. The surge
levels applied in these tests will be different for these two areas. The AC area may
be classified as class 3 or 4 in EN 61000-4-5 and voltage levels defined by standards
for the test signals are representative of perturbations in AC power distributions
accomplishing the system integration in a safe way.
In the DC area the signal level to be applied during tests is not completely definedby standards but must be defined by each sub-system itself. Assuming thetransients in the AC system are highly attenuated by the AC protection system andthey do not propagate into the DC line, transients in the DC area are defined by thelevel of energy involved by the FEE and cable bundle granularity. Some preliminary
studies have been conducted to quantify the energy level involved in thesetransients and to relate them to design considerations as granularity, type ofprotections, etc. Studies presented in this chapter concluded that good design
considerations such as cables bundles grouped per power levels, small granularity,
voltage regulators and current limiters, lead to easy and feasible solutions to protect
the FEE against surge transients. Bad topologies may lead to difficult solutions toprotect the system and can result in a potential problem during the normal
operation of the detector. This research has shown that the best solution to protect
the system is using the technique of protection per stages, which helps to deal withthe limitations of real protection components. Each component of the protection
stage attenuates partially the transient generated in the system. Transient voltage
suppressors (TVS), capacitors and voltage regulators should be included in both the
distribution system and the sensitive equipment connected to it. Proper
coordination among these devices is necessary to divert the transient energy
through the device with suitable rating. In addition when TVS are used,
coordination with over-current protection should be evaluated. In low-voltage
applications, the clearing time of fuses is affected by the connection of TVS at the
distribution bus.
Fast transients studies are only presented but not studied in detail because this typeof transients may occur in CMS in the AC distribution system and it is fully coveredby the standard EN 61000-4-4.
It is not yet possible to fix the level and waveform for the detector as the final designof the power supply system is not defined. As a first approximation, the selectedlevel and waveform may be based on the area where the equipment under test will
be placed in CMS (AC area & DC area). In the DC area, the test level may be
specified per sub-subsystem because the bundles of cables and protections are
divided per sub-system. These levels will not be only based on the preliminary
studies shown before. New simulations and test of the sub-detector electronics have276

7.7. References

to be performed to get a statistical data about the type and the number of transients
occurring in this kind of experiments.

7.7 References[1] IEEE-Std. 1100-1999, "IEEE Recommended practice for powering and grounding
electronic equipment", Ed. 1999.
[2] T. Gruzs, "Design Considerations for Powering 415-Hz Computer Systems", IEEE
Transactions on Industry Applications, Vol. 25, N-6, pp 1102-1109 November December 1989.
[3] Keith Armstrong, "Design Techniques part 6 - ESD, Dips, Flicker, Dropouts
electromechanical switching and factor correction", EMC & Compliance Journal,
http://www.compliance-club.com/.
[4] Keith Armstrong, "EMC Testing Part 3- Fast Transient Burst, Surge, Electrostatic
Discharge", EMC & Compliance Journal, http://www.compliance-club.com/.
[5] International Standard-IEC-61000-4-11 "Electromagnetic compatibility (EMC) Testing and measurement techniques- Voltage dips, short interruptions and voltage
variations immunity test", Basic EMC publication (Ed. 1995).
[6] C.R.Paul, "Introduction to Electromagnetic Compatibility", Wiley-Interscience, 1992,
ISBN - 0 - 471-54927-4.
[7] Tim Williams, "EMC for product designers", Paraninfo 1997 ISBN-0-12-55710-0[8] W. Rhoades, "Congruence of low voltage power main transient designs", Proc.
National Symposium on Electromagnetic Compatibility, pp- 285 -293, May 1989
[9] International Standard - IEC - 61000-4-5, "Electromagnetic compatibility (EMC)Testing and measurement techniques-surge immunity test", Basic EMC publication (Ed.
2001)
[10] R. D’Alessandro, G. Parrini, "General guidelines for the tracker power supply
system", Private communication.
[11] R. Hammarstrom, "Mutual wire inductance in MSC prototype", Private
communication.
[12] A. Ciccolella, F.G. Canavero, "Stochastic prediction of wire coupling interference",
Proc. IEEE International Symposium on Electromagnetic Compatibility, pp-51 -56,
August 1995.

277

7. Susceptibility to transient perturbation[13] G. T. Capraro, C. R. Paul, "A probabilistic approach to wire coupling interference
prediction", Proc. 4th International Zurich Symposium on Electromagnetic
Compatibility, Zurich, Switzerland, pp. 267-272, February 1981.
[14] C. R. Paul, "Sensitivity of crosstalk to variation in wire position in cable bundles",Proc. 7th International Zurich Symposium on Electromagnetic Compatibility,
Zurich, Switzerland, pp. 617-622, March 1987.
[15] T. Robbins, "Fuse model for over-current protection simulation of DC distribution
systems", Proc. International Telecommunications Energy Conference, INTELEC ’93,
Vol 2, pp 336-340, September 1993.
[16] R. B. Standler, "Transmission line models for coordination of surge-protective
devices", Proc. IEEE International Symposium on Electromagnetic Compatibility,
Dallas, USA, pp 363-368, August 2003.
[17] E.P. Dick
R.W. Cheung,
J.W. Porter, "Generator models for over voltage
simulations", IEEE Transactions on Power Delivery, Vol 6, Issue 2, pp 728-735, April
1991.
[18] R. Siegert, O. Mohammed, "Evaluation of a hybrid surge testing generator
configuration using computer based simulations", Proc. IEEE Southeastcon '99,
Lexington, USA, pp 193-196, 1999.
[19] F.A. Fisher, F.D. Martzloff, "Transient control level philosophy and implementation",
Proc. International Symposium on Electromagnetic Compatibility, Montreux, 1977.
[20] R.B. Standler, "Standard Waveforms for Surge Testing: Experimental Evaluation and
Proposed New Criteria for Tolerances", Proc. International Symposium on
Electromagnetic Compatibility, Zurich, pp 115-124, March 1989[21] F.D. Martzloff, "The development of a guide on surge voltages in low-voltage AC
power circuits", Application note 9769, Littlefuse Company.
[22] ANSI/IEEE C62.41-1980, "Guide on Surge Voltages in AC Power Circuits Rated up
to 600V ", IEEE publication (Ed. 1980)[23] F. Martzloff, T.F. Leedy, "Electrical fast-transient tests: applications and limitations",
IEEE Transactions on Industry Applications, Vol 26, Issue 1, pp 151-159, January
February 1990.
[24] International Standard - IEC - 61000-4-4, "Electromagnetic compatibility (EMC) Testing and measurement techniques-Electrical fast transient/burst immunity test", Basic
EMC publication (Ed. 1995).
[25] B. Cure, "CMS Coil Powering Circuit - Safety Issues", Engineering design review
EDR-MAG-02, slides, 29-30 March 2001.

278

Conclusions

The EMC study of a high-energy physics detector has been presented. This analysis
is fundamental to ensure the integration and fully functionality of the electronicsystem of the detector. The main goal of this study has been to evaluate EMC
problems at system level. The main topics analysed in this thesis have been:• The grounding• The conducted emissions and filtering of the PS• The noise propagation effects on cables• Immunity of the FEE-

Immunity at design stage.

-

Immunity test of the FEE• Transients on the PS distribution system
The main results and conclusions obtained from the analysis are detailed below:
1. EMC studies have started with the analysis of grounding and shielding issues
that are the baseline for a good EMC policy. Based on electric safety, equipment
protection and equipment performance, the main rules to be applied to the CMS
detector have been determined.
2. Conducted emissions can affect the performance of the CMS detector. For thatreason, it is necessary to conduct normalized tests (close to aerospace EMI rules andmilitary standards) in order to measure the conductive emission level at the output
and input of the power supplies. In some cases, EMI filters are needed to attenuatethese emissions to acceptable levels.
• The filter selection is based on its insertion loss, which strongly depends on theexternal impedance connected to its terminals. Therefore, it becomes necessary
to define adequate load and source impedances that allow selecting a robust
filter to achieve the low emission levels imposed.
• Input filters interact adversely with DC-DC converters at low frequency,
resulting in severe performance degradation or even instability. The instability
limits the number of DC-DC converters to be connected to the same power
distribution bus. This problem can be controlled defining the number of units
connected to the same bus and the damping.

279

Conclusions

3. Long power cables installed in the detector produce undesirable effects on thenoise propagation between the power supply (PS) units and the front-end
electronics (FEE). These effects must be seriously considered as they strongly affectthe performance of the CMS detector:• For unbalanced loads, which are typical in high-energy physics, both CM andDM mode signals are amplified at the resonance frequencies of the cable. In
particular, when a capacitor filter is included across the load terminals, the CM
and DM currents flowing through the cable are amplified, but the DM and CM
voltage across the load terminals are effectively attenuated. This input capacitor
has the additional effect of decreasing the CM to DM and DM to CM conversion
and reducing the amount of noise that can affect the performance of the FEE.
• FEE input filter with low impedance (capacitive input) amplifies CM currents
from the power supplies, leading to severe radiation from the cable. To attenuate
this effect, it is highly recommended to use shielded power cables. Moreover,this shield contributes to keep these noise currents away from the detector
ground, reducing the interaction among sub-systems. The use of non-shieldedcables can force to use power supplies with a very low conducted CM emission
level to decrease the amount of noise radiated by the cables.
• For external shield currents, the factor that determines the amount of noise
coupled to the central conductors is defined by the surface transfer impedance ofshielded cable. The use of copper braid shields helps to decrease this coupling
and therefore decreases the amount of noise affecting the performance of theFEE.
4. The susceptibility of the FEE is the most important part of the EMC studies. The
susceptibility should be dealt at the design stage, where by means of simulations, it
is possible to select the best FEE topology. As example, some studies have been
carried out in the Forward HCAL-FEE, whose main conclusions are:
• The topology of the connection between the FEE and the detector presents
several limitations to common mode rejection and radiated noise immunity if
this one is implemented with separated coaxial cables. The rejection to spurious
signals can be improved by selecting two coaxial cables with a single braided
shield.
• The use of a ground resistor to isolate the HV to the input circuit of the FEE,
improves the common mode rejection of the system proportionally to its
magnitude, decreasing at high frequency. However, it has a negligible effect on
the noise generated by external electromagnetic fields. This difference between
both cases can be explained by the different nature of the circuits.
5. EMI tests are carried out to evaluate the susceptibility of the FEE. These tests
allow taking the necessary corrective action in case of low noise immunity, as wellas, determining the noise level required at the power supply outputs. Tests applied
on a prototype of HCAL FEE have drawn the following main conclusions:• The principal cause of the FEE performance degradation is the CM current. CM
280

Conclusions

currents are coupled into the sensitive path of the FEE by currents flowing
through the shield, by the DM to CM conversion and by CM current generated
by the power supplies. The DM currents contribute to the degradation of the
FEE mainly through DM to CM conversion.
• In order to improve the immunity of the FEE to noise currents, there are several
options such as: the selection of good grounding connections and the use of CM
filters and shielded power cables.
• The immunity curves of the system can be used to define a new noise level
required at the output of the power supply. For HCAL FEE this level results very
different from the level defined by the European standards. The definition of astandardized noise level at the output of the power supplies without theknowledge of the immunity curves of the system can lead to problems. It is
important the definition of this level for all the sub-systems of the detector. Later,
based on this curves, it is possible to define the emission level for the power
supplies or select the European, American or Military standards that fits betterto the requirements of the detector. In any case, the standardized level should be
extended above the 30MHz limit because the FEE is very sensitive in thatfrequency range.
6. For transient perturbations in the power distribution system, the CMS complex
can be divided in two areas. The first one corresponds to the AC distribution(between the motor-generators and the counting room / the periphery of thedetector) and the other the LV-DC distribution system (between the periphery of thedetector and the FEE). Each of them is perturbed by transient voltages withdifferent characteristics, making necessary to analyze this problem separately.
• The AC power supply distribution is well defined by the standards. According
to these standards, all the equipment placed in the AC area can be classified aseither class 3 or class 4 for surge transients and class 3 for burst. Tests andtransient levels to be applied to measure the immunity to surges, bursts andvoltage dips of the equipment are well established.
• Transients generated in the DC distribution system itself are the only one able toaffect the FEE in the detector because the transients generated in the AC line areattenuated at the input of the DC system.
• The limits for the transients in the detector area depend on the front-end
electronics immunity. From the measurements performed in a prototype of
HCAL-FEE it is possible to define the baseline of future EMC policy of thedetector.
-

Voltage dips tests applied to digital power cables showed that the digital partof the HCAL-FEE is very susceptible to these transients. Dips of a few volts(around 3 volts) and short duration (around 10ms) can produce a fail in theclock signal, being necessary to reset the whole system to recover. Voltage
dips applied to the analogue line of the HCAL-FEE induced a temporary gain
loss of the amplifier.

-

Surge tests have been focused on the characterization of the transient

281

Conclusions

protection devices included at the input of the HCAL-FEE and the effect of
the applied signal in the FEE. These tests showed that the best solution toprotect the FEE is using a protection system based on protection by stages. In
this topology, each component of the protection chain attenuates partially thetransient generated in the system. At HCAL we studied the combination oftransient voltage suppressors, capacitors and voltage regulators to suppress
the effect of the transient on the FEE bias voltage.
-

Fast transients are not considered in detail as it is very unlikely to be present
in this area of the CMS detector.

As a general statement, these studies conducted in the CMS detector show that thefirst step in the EMI control is the selection of good topologies for the threecomponents of the system, the FEE, the PS and the cables. The former increases theimmunity of the FEE, the second decreases the emission of the PS to acceptablelevels and the third one is used to control the coupling paths. The selection of only
one of these components as the only component oriented to cancel the EMI issues isa tremendous mistake. The system should be thought as an unique identity inwhich each component has a specific task, being the combination of all of them thefinal solution to ensure a good detector integration.

282

Thesis contribution and future work

This thesis has clearly fulfilled the proposed objectives:• Description of the EMC environment of CMS and quantification of noise level
and immunity of some sub-systems.
• Pre-definition of emission and immunity specifications for noise level.
• Definition of normalized tests for the equipment to comply with those levels.
The main contribution of this work has been the development of a methodology toface the EMC aspects of a HEP experiment following of a systematic approach. The
thesis has created the baseline for the CMS EMC policy, which can be extended toother experiments. This is the first time this type of proposal is presented in a HEP
environment.
The definition of the susceptibility to conducted noise and transients for the rest of
the sub-systems will allow having all immunity levels of the detector. As a future
work it is intended to implement this plan for the rest of the CMS sub-systems. This
thesis opens several work lines that should be followed to define the future EMC
policy in CMS. The EMC plan for CMS will have four basic steps that should be
followed for all the sub-systems located in the AC and DC areas of the detector:• Power supply/distribution block diagram• Grounding Scheme• Immunity tests-

RF conducted noise (Common mode and Differential mode)-Harmonics Tests• Emission tests-

Immunity to radio frequency disturbances test.

-

Voltage dips, short interruptions and voltage variations immunity test.

-

Surge immunity test - Over-voltage-

Burst immunity test

Basically this thesis presents a global analysis of EMC at macro-system level. A
detailed analysis of every specific EMC issue like power supply emissions, filter
design and cables effects is out of the scope of this work and constitutes by itself asubject of research. In this sense, this thesis opens several work-lines, which should283

Thesis contribution and future work

be followed in future studies if a general EMC policy wants to be addressed in the
HEP community as it has been done in other communities such as aerospace,telecommunications and automobile. The main work-lines necessary are listed
bellow, some of them constitute a subject to develop a new thesis.
• Data collection and characterization of immunity level of the FEE and theelectronic equipment installed in HEP experiments.
• Characterization and parameterization (level, waveform, energy) of transient
perturbations likely to be present in any experiment in HEP.
• Definitions of incompatibilities between the experiments and the accelerator.
• Study of EMI in the detector front-end electronics. EMI design oriented of the
FEE.
• Noise and transient analysis in cable bundles.
• Development of protection devices for very low DC power voltage distribution
in hostile areas, analysis of risk assessment.
• EMC impact in the upgrade of the detectors.

284

Conclusiones - Versión en castellano

En este trabajo se presenta el análisis de compatibilidad electromagnética en un
detector de física de altas energías. Este estudio es fundamental para asegurar laintegración y total funcionalidad de la electrónica del detector. La tarea principal deeste estudio ha sido evaluar la problemática de EMC a nivel de sistema. Los temas
principales analizados en la tesis han sido los siguientes:• La puesta a tierra• Emisiones conducidas y filtrado de las fuentes de alimentación• Los efectos de la propagación de ruido en los cables• Inmunidad del FEE-

Inmunidad en la fase de diseño-

Ensayos de inmunidad del FEE• Transitorios en el sistema de potencia
Los principales resultados y conclusiones del análisis de estos temas se detallan acontinuación:
1. Los estudios de EMC comenzaron con el análisis de la puesta a tierra y blindaje
de los equipos del experimento. Estos factores determinan la base de una buena
política de EMC. Una buena puesta a tierra y un blindaje óptimo se han de basar
siempre en especificaciones de seguridad eléctrica, protección de los equipos yóptimo funcionamiento del sistema. En este primer análisis se detallan las
principales características y recomendaciones válidas para el caso particular de
CMS.
2. Las emisiones conducidas pueden afectar al buen funcionamiento del detector.
Por esta razón, es necesario definir una serie de ensayos normalizados (similares aestándares aeroespaciales y militares) con el propósito de estimar el nivel de ruido
exigido a la salida y a la entrada de las fuentes de alimentación. En algunos casos
es fundamental el uso de un filtro de EMC con el fin de disminuir las emisiones aniveles aceptables para el detector.
• La selección de los filtros está basada en sus propiedades de funcionamiento yrendimiento, las cuales suelen venir definidas en función de sus perdidas por
inserción. Estas últimas dependen enormemente de las impedancias que
aparecen en los terminales del filtro. Por lo tanto, es necesario definir las
impedancias de carga y fuente adecuadas que permitan seleccionar el filtro más

285

Conclusiones - Versión en castellano

adecuado a las necesidades o requerimientos del sistema, manteniendosuficiente margen de seguridad.
• Los filtros de entrada pueden interferir con los convertidotes DC-DC en baja
frecuencia. Como resultado de esta interacción se produce una brusca
degradación del funcionamiento del sistema que puede dar lugar ainestabilidades. Este problema limita enormemente el numero de unidades
conectadas a una misma fuente primaria, que en el caso de CMS es unconvertidor de alterna a continua. Este problema puede ser sencillamente
controlado definiendo el número de unidades conectadas a la misma fuente
primaria.
3. La propagación del ruido a través de los cables largos que unen las fuentes dealimentación y el FEE produce efectos no deseados que pueden degradar elcorrecto funcionamiento del detector. Por tanto, estos efectos deben ser tomados en
consideración.
• Para el caso de cargas no equilibradas, muy típicas en física de altas energías,tanto las señales de modo común como las de modo diferencial se amplifican
por efecto de las resonancias del cable. Concretamente, al colocar uncondensador a la entrada del FEE, las corrientes de modo común y mododiferencial son amplificadas, mientras que, por contra, las tensiones generadas
por estos modos en el terminal de entrada son atenuadas. Además, este
condensador presenta un efecto adicional beneficioso, que consiste en ladisminución de la conversión de modo común a modo diferencial, con lo que
disminuye la cantidad de ruido que pudiera afectar al sistema.
• Sin embargo, el uso de este filtro capacitivo a la entrada del FEE puede conducira una elevada radiación del cable, generada por el modo común inyectado en los
cables por parte de las fuentes de alimentación. Con el fin de atenuar este efecto,
se recomienda el uso de cables de potencia blindados. El blindaje de estos cables
contribuye, por otra parte, a mantener la estructura de referencia del detector, asícomo otros sistemas, libres de estas corrientes de ruido. El uso de cables sin
blindaje podría forzar el uso de fuentes de alimentación con niveles de ruido a lasalida extremadamente pequeños, como única solución que mantenga bajo elefecto de radiación de los cables.
• Respecto a las corrientes externas del blindaje, el principal factor que determina
la cantidad de ruido que se puede acoplar en los conductores centrales estádefinida por la impedancia superficial de transferencia del blindaje. El uso deblindaje del tipo entrelazado de cobre (copper braids) ayuda a disminuir lacantidad de ruido acoplado al sistema, mejorando el rendimiento de laelectrónica.
4. La susceptibilidad del FEE constituye uno de los elementos más importantes en
este estudio de EMC. La susceptibilidad debe ser analizada de antemano en la fasede diseño, donde mediante simulaciones, se puede seleccionar la mejor topologíadel FEE desde el punto de vista de EMC. Como aplicación de este tipo de estudios,286

Conclusiones - Versión en castellano

se ha realizado un análisis en fase de diseño del conjunto electrónica-detector del
Forward Hadron Calorimeter, cuyas principales conclusiones son las siguientes:• La topología usada en dicha conexión limita enormemente la capacidad delsistema para rechazar ruido de modo común y disminuye la inmunidad delconjunto al ruido radiado. El origen de esta problemática se encuentra en el uso
de cables coaxiales separados, pudiendo solventarse en parte este problema
usando un cable doble coaxial con un único blindaje de cobre común.
• El uso de una resistencia para aislar la componente de señal de la parte de altatensión del detector mejora el rechazo del sistema al
modo común
proporcionalmente con el valor de esta resistencia, disminuyendo este efecto aalta frecuencia. Sin embargo, dicho efecto es despreciable en caso de que elruido acoplado al sistema sea generado por campos externos. La diferencia entre
ambos casos viene motivada por la diferente naturaleza de los circuitos
implicados en ambos casos.
5. En una segunda fase del análisis de la susceptibilidad del FEE, se han realizado
experimentos con el objetivo de definir la inmunidad del FEE al ruido conducido.
Estos ensayos permiten evaluar la inmunidad de la topología aplicada en el FEE,
tomar las medidas correctoras en el caso de presentar una baja inmunidad y
determinar el nivel de ruido que será necesario especificar a la salida de las fuentes
de alimentación. Una serie de ensayos de inmunidad que se realizaron en un
prototipo del FEE han permitido obtener las siguientes conclusiones:• Las corrientes de modo común son las principales responsables de la
degradación del funcionamiento del FEE. Estas corrientes normalmente son
generadas por corrientes de blindaje, por las fuentes de alimentación, así comopor la conversión de modo diferencial a modo común.
• Existen diversas acciones que se pueden realizar con el fin de mejorar la
inmunidad del FEE, como por ejemplo elegir muy buenas conexiones al planode referencia, instalar filtros de modo común a la entrada de la electrónica y usar
cables blindados de potencia.
• Las curvas de inmunidad del sistema se pueden usar para definir los nuevos
niveles de emisión de ruido requeridos a la salida de las fuentes de alimentación.
En el caso de HCAL-FEE estos niveles resultaron ser muy diferentes de losfijados por los estándares europeos. Esto lleva a concluir que la definición de unnivel de ruido basado en estándares sin el conocimiento de las características deinmunidad del FEE puede llevar a serios problemas. Sin embargo, si estos
niveles de inmunidad son definidos, es posible que, basándose en ellos, se pueda
elegir el estándar europeo, militar o espacial, que mejor se ajuste a lasnecesidades de nuestro sistema. En todo caso, el rango de frecuencias ha de ser
extendido por encima de los 30 MHz, dado que el FEE es muy sensible en ese
rango de frecuencias.
6. En el caso del análisis de transitorios, el sistema de distribución de CMS está
dividido en dos partes principales. Una que corresponde al sistema de distribución287

Conclusiones - Versión en castellano

trifásica del detector (Generadores/Counting room - periferia del detector) y la otra
que corresponde al sistema de distribución de corriente continua entre la periferiadel detector y el FEE. Cada uno de ellos presenta transitorios de diferentes
características, resultando necesario analizar el problema de los transitorios deforma separada.
• El sistema de distribución trifásica está perfectamente definido en losestándares. Los ensayos a aplicar para definir la inmunidad de los equipos
contra sobre-tensiones, transitorios rápidos o caídas de tensión temporales están
perfectamente definidos por los estándares. De acuerdo a estos estándares, todo
equipo localizado en esta área se puede clasificar, tanto como clase 3 o clase 4,
para el caso de transitorios lentos y como clase 3 para transitorios rápidos.
• Los transitorios generados en el sistema de distribución de continua sonúnicamente aquellos generados por el mismo sistema ya que los transitorios
generados en el sistema de AC son atenuados a la entrada del sistema decontinua.
• Los límites de los transitorios en el área del detector dependen únicamente de las
características del sistema de distribución de esta área. Una serie de medidas y
ensayos realizados en un prototipo del HCAL-FEE han permitido establecer una
pautas que pueden ser usadas para fijar la base de la futura política deprotección del detector contra los transitorios.
-

Ensayos de pérdida de tensión muestran que la parte digital del HCAL-FEE
es la más sensible a este tipo de transitorios. Pérdidas de tensión de unos 3
voltios durante cortos periodos de tiempo, unos 10 ms, pueden producir
fallos en la señal de sincronización que provoque un fallo general en todo elsistema, resultando necesario reinicializar el detector. En el caso de aplicar lapérdida de tensión a la línea analógica del HCAL-FEE tan sólo se produce
una perdida temporal de la ganancia del amplificador.

-

Con los ensayos y análisis de sobre tensión en esta área se ha tratado decaracterizar el sistema de protección necesario a la entrada del FEE. Los
ensayos mostraron que el mejor sistema de protección es aquél basado en una
protección por etapas. Cada componente de este sistema de protección
atenúa parcialmente el transitorio generado en el sistema. El sub-sistema
HCAL ha estudiado el uso de supresores de transitorios, condensadores yreguladores de tensión como cadena de protección del FEE contra lostransitorios.

-

Los transitorios rápidos no son considerados en detalle al no estar tan claro
su presencia en esta área del detector.

Como resumen general de la tesis hay que indicar, que estos estudios realizados
para CMS muestran que la primera etapa en el control de perturbaciones
electromagnéticas conducidas en el detector es una buena elección de las topologíasde los tres principales componentes del sistema de distribución, FEE, fuentes ycables. La primera incrementa la inmunidad del detector, la segunda disminuye las288Conclusiones - Versión en castellano

emisiones de las fuentes de alimentación a niveles aceptables y la tercera es usadapara controlar los caminos de acoplamientos entre componentes. La selección deuno de estos componentes como único responsable de la cancelación de problemasde interferencias electromagnéticas es un enorme error. El sistema debe
considerarse como una única identidad en la que cada componente tiene una tarea
especifica, siendo la combinación de todos ellos la solución final que asegura lacorrecta integración del detector.

289

Conclusiones - Versión en castellano

290

Objetivos de la tesis & Futuros
Trabajos -Versión en castellano

Esta tesis ha cumplido claramente los objetivos propuestos con el fin de abordar los
problemas de interferencias electromagnéticas en el detector.
• Se ha descrito el ambiente electromagnético del detector• Las especificaciones de emisiones e inmunidad al ruido en CMS han sido
pre-definidas.
• Se han definido los ensayos normalizados que permitan integrar el detector deforma segura.
La principal contribución de esta tesis ha sido el desarrollo de una nueva
metodología que permita hacer frente a cuestiones de EMC en experimentos defísica de altas energías mediante un análisis sistemático de los fenómenos
electromagnéticos dentro del detector. Esta tesis será tomada como base para
desarrollar la futura normativa a seguir para integrar el detector de forma segura.
Dicha metodología puede ser perfectamente aplicada a otros experimentos. Esta es
la primera vez que este tipo de análisis y propuesta es presentado en física de altas
energías.
La definición de la susceptibilidad al ruido conducido y a transitorios del resto delos subsistemas de CMS permitirán caracterizar el nivel de inmunidad del detector
y finalmente establecer el nivel requerido por CMS. Como futuro trabajo seprocederá a implementar un plan de EMC en el resto de subsistemas de CMS. El
plan para CMS tendrá 4 etapas básicas, que deberán aplicarse a cada subsistema.
• Diagrama del sistema de distribución de potencia• Esquema de la puesta a tierra de los equipos y electrónica.
• Ensayos de inmunidad-

A ruido de radio frecuencia conducido-

Ensayos de armónicos• Ensayos de inmunidad-

Inmunidad a ruido conducido de radio frecuencia-

Inmunidad a perdidas de tensión-

Inmunidad a transitorios lentos

291

Objetivos de la tesis & Futuros Trabajos -Versión en castellano-

Inmunidad a transitorios rápidos

Esta tesis ha intentado hacer frente a la problemática de EMC en física de altas
energías, y más en concreto en el detector CMS. Básicamente presenta el problemadel análisis global de EMC en el ámbito de macro sistema. Un análisis detallado decada tema de EMC como emisiones de las fuentes de alimentación, diseño de filtros
y efectos en los cables está fuera de los objetivos y el alcance de este estudio,
constituyendo cada uno de ellos en sí mismo un tema de investigación. En este
sentido, esta tesis abre diferentes líneas de trabajo que deben ser estudiadas en
detalle si se quiere aplicar una normativa general para el mundo de física de altas
energías, tal como han hecho otras comunidades, como la industria del automóvil,la industria aeroespacial, telecomunicaciones y en general toda familia de equipos
electrónicos que se puedan encontrar en el mercado. Las principales líneas detrabajo necesarias para desarrollar los futuros estándares de EMC en física de altas
energías se enumeran a continuación. Algunas de ellas pueden constituir en sí
mismas un tema de tesis.
• Caracterización de la inmunidad de cualquier FEE y equipo electrónico
instalado en un experimento de física de altas energías.
• Caracterización y parametrización de los tipos de transitorios (nivel, forma de
onda, energía) que puedan presentarse en un detector de física de altas energías.
• Definición de incompatibilidades entre el acelerador y el detector.
• Estudios de EMC el FEE del detector. Diseño de FEE orientado a EMC.
• Análisis de ruido y transitorios en agrupaciones de cables.
• Desarrollo de sistemas de protección para distribuciones de potencia a muy baja
tensión en ambientes hostiles. Análisis y evaluación de riesgos.
• Impacto de aspectos de EMC en la actualización del acelerador.

292

Appendix A - MTL
A.1 Internal impedances - Skin effect
The DC resistance of a conductor is computed by the typical equation.

ρ⋅l
R DC = --------A(A.1)

As the frequency increases, the current over the wire cross section tends to flowcloser to the outer part of the conductor, producing an increase in the resistance ofthe conductor. This is known as the skin effect. For well developed skin effect thecurrent can be considered to be uniform distributed over an annulus at the surfaceof the wire of depth, a skin depth δ .

1
δ = -------------------------------π ⋅ f ⋅ µo ⋅ σ(A.2)

On the other hand, the other component of conductor, which is affected by the skineffect, is the internal inductance. The internal inductance is the inductance arising
due to the flux linkage inside the conductor for its own current. This values is quite
small if it is compared with the self or mutual inductance of a conductor, and for thenhcase of circular conductor is equal to l i = 50 ------ . In the case of internal inductance,mthe skin effect produces a decrease of the value of the internal inductance up to acertain frequency in which this value can be considered negligible.
The increase of the wire resistance and the diminution of the inductance for the
HCAL and Tracker power are shown in figures 4.8,4.9,4.14 and 4.15.
To model this cable and include these effects in the frequency domain, theimpedance of the conductor is divided in two parts; the internal and the external.
The former corresponds to the values of the external inductance of the conductors(which are constant), and corresponds to the values of the conductor inductance athigh frequency. The second to the values of the conductor resistance and internal
inductance, which depends on the skin effect. This impedance can modeled infrequency domain as [1][2]:293

Appendix Af
Z = RDC + R DC ⋅ ---- ⋅ ( 1 + j )
fo

(A.3)

r
The value of f o corresponds to the frequency where δ = --- .
2

A.2 Characteristic impedance
The definition of the characteristic impedance of a multi-conductor transmission
line involves the definition of a matrix. The concept is similar to the case of a single
conductor line, but mathematically the calculation is more involved.
Based on equations 4.35 and 4.56, it may be obtained that:
–1
–1
ˆ
ˆ ˆˆ+
– γz
γz
Vˆ ( z ) = ( Yˆ ⋅ Tˆ ⋅ γ̂ ⋅ Tˆ ) ⋅ Tˆ ⋅ ( e ⋅ I m + e ⋅ I m )(A.4)

Under these considerations the characteristic impedance matrix is defined as:–1–1Zˆ o = Zˆ ⋅ ( Tˆ ⋅ γ̂ ⋅ Tˆ )(A.5)

For a single transmission line the parameters Z and γ̂ are scalar and Tˆ = 1 . In
this case, the definition of the characteristic impedance is.
–1Zˆ o = Zˆ ⋅ ( γ̂ )γ̂ =Zˆ ⋅ Yˆ =(r + j ⋅ ω ⋅ L) ⋅ (j ⋅ ω ⋅ C) = α + j ⋅ β

(A.6)

(A.7)

where α is the attenuation constant and β is the phase constant per unit length.
From equation A.6 and A.7, the final expression of the characteristic impedance fora single transmission line is obtained as:

Zo =r+j⋅ω⋅L-------------------------j⋅ω⋅C(A.8)

For the multi-conductor case, the characteristic impedance given by the expression
A.5 is numerically evaluated. It corresponds to the HCAL power cable and theparameters are measured1 at 1 MHz. The characteristic impedance matrix in Ω is:Zˆ o = 25.81 4.542 Ω
4.542 26.07
1 See chapter 4 - section 4.3
294

(A.9)

A.2. Characteristic impedance

A direct measurement of this characteristic impedance is possible similarly to thecase of single conductors using a time domain reflectemeter (TDR). Figure A.1
shows a TDR implemented in Pspice where a pulse generator of 1 A is applied
between one line and the shield. The other side of cables is left floating as well as
the other conductors. The voltage of the first incident wave is measured at thesending end for all conductors. This magnitude is coincident with the first row of
the characteristic impedance matrix. Figure A.2 shows the results of simulation for
HCAL power cable.

Figure A.1: TDR implemented in Pspice to measure the characteristic impedance matrix.

30V

25V

20V

Z11=25.81
15V

Z12=4.54
10V

5V

0V
0s

1ns
V(Black:A+)-V(Ref:1)

2ns

3ns
V(Blue:A+)-V(Ref:1)

4ns

5ns

6ns

7ns

8ns

9ns

10ns

Time

Figure A.2: Characteristic impedance matrix - First row.

295

Appendix A

A.2.1

CM and DM characteristic impedance matrix

For a pair of conductors, it is possible to decompose the currents in each wire I 1and I 2 in a set of orthogonal current components. One of these components iscalled differential mode current and it is represented by two equal currents flowing
through the cables in opposite direction. The other component is called common
mode current and it is represented by identical currents and flowing in the same
direction. This decomposition is shown in figure A.3 and defined by equations A.10
and A.11.

I CM

I1

I DM

I2
Figure A.3: CM and DM currents definition.

I 1 = I cm + I dm

(A.10)

I 2 = I cm – I dm

(A.11)

which can be expressed in matrix form as:

I1
I2

= W⋅

I cm

(A.12)

I dm

where W represent the transformation matrix from the normal mode ofpropagation to the CM and DM of propagation:

W = 1 1
1 –1

(A.13)

This equation is a nonsingular transformation between both modes of propagation.
Therefore, the CM and DM currents can be calculated by:

I cm
I dm

296

I
1
= --- ⋅ 1 1 ⋅ 1
2 1 –1
I
2(A.14)

A.2. Characteristic impedance

Based on the definition of both modes of propagation, it is possible to define a
DM-CM characteristic impedance matrix applying a change of reference:–1Zˆ ( CM – DM )O = ( W ⋅ Zˆ o ⋅ W ) =30.4 –0.13 Ω– 0.13 21.4

(A.15)

As it has been developed before, a direct measurement of this characteristic
impedance is possible similarly to the case of single conductors using a time
domain reflectemeter (TDR). Figure A.4, shows a TDR implemented in Pspicewhere two pulse generator of 1 A are connected between one line and the other in
CM configuration.

Figure A.4: TDR implemented in Pspice to measured the CM-DM characteristic
impedance matrix - CM configuration.
The voltage of the first incident wave is measured at the sending end for all
conductors and the transformation shown below is applied to calculate the CM andDM voltages:

V1 – V2
V dm = -----------------2(A.16)

V 1 + V2
V cm = -----------------2(A.17)

These magnitudes are coincident with the first row of the CM-DM characteristic
impedance matrix. Figure A.5 shows the results of simulation of HCAL power
cable.

297

Appendix A

40V

30V

20V

Z12=-0.13
Z11=30.4
10V

0V

-5V
0s

1ns
2ns
(V(Blue:A+)-V(Black:A+))/2

3ns
4ns(V(Blue:A+)+V(Black:A+))/2

5ns

6ns

7ns

8ns

9ns

10ns

Time

Figure A.5: CM-DM characteristic impedance matrix - First row.
Figure A.6a, shows a TDR implemented in Pspice in DM configuration. For thisconfiguration a pulse generator of 1 A is connected between one line and the other.
The voltage of the first incident wave is measured at the sending end for all
conductors and the previous transformation is applied again. This magnitude iscoincident with the second row of the characteristic impedance matrix. Figure A.6bshows the results of simulation of HCAL power cable.

A.6a

30V

A.6b

20V

Z22=21.4
10V

Z21=-0.13

0V

-5V
0s

1ns
2ns
(V(Blue:A+)+V(Black:A+))/2

3ns
4ns
5ns
(V(Blue:A+)-V(Black:A+))/2
Time

6ns

7ns

8ns

9ns

10ns

Figure A.6: TDR implemented in Pspice to measured the CM-DM characteristic
impedance matrix - DM configuration - CM-DM characteristic impedance matrix - Second
row.

298

A.3. MTL programs validation

A.3 MTL programs validation
As an example to validate the MATLAB program used in previous studies, two
examples of multi-conductor system shown in [2] have been studied and the resultsof the MATLAB program are compared with the values measured and calculated.

A.3.1 Ribbon cableThis example consists on the study of the frequency domain solution of a threeribbon cables connected as it is shown in figure A.7.

Figure A.7: Circuit under study.
The terminal configurations, characterized as a generalized Thevenin equivalent
structures are:Vˆ s = 0
1(A.18)Vˆ L = 00(A.19)Zˆ L = 50 0 Ω
0 0(A.20)Zˆ S = 50 0 Ω
0 50(A.21)The total length of the line is 2 meters and the per unit length parameters for thisconfiguration are:299

Appendix AµH
L = 0.7485 0.5077 -------0.5077 1.0154 m(A.22)

37.432 – 18.716 pF------– 18.716 24.982 m(A.23)C =The resistance skin effect has been included by determining the frequency where
the radius of one of the cables are equal to two skin depths, which according to theequation of A.2, where fo is equal to 4.332 MHz.
⎛⎛⎜⎜2 ⋅ 0.19444 ⋅ ⎜ 1 +
⎜
⎜
⎝
⎝

⎞⎞
⎟
f
---------------------------------------⎟ ⎟⎟⎛ 4.332 ×106⎞ ⎟
⎠
⎝
⎠ ⎠

R =
⎛
⎞
⎜
⎟
f
0.19444 ⋅ ⎜ 1 + ---------------------------------------⎟⎜⎛ 4.332 ×106⎞ ⎟⎠
⎝
⎝
⎠

⎛
⎞
⎜
⎟
f
0.19444 ⋅ ⎜ 1 + ---------------------------------------⎟⎜⎛ 4.332 ×106⎞ ⎟
⎝
⎝
⎠⎠
⎛
⎛
⎜
⎜
2 ⋅ 0.19444 ⋅ ⎜ 1 +
⎜
⎜
⎝
⎝

⎞⎞
⎟
f
---------------------------------------⎟ ⎟⎟⎛ 4.332 ×106⎞ ⎟⎠
⎝
⎠ ⎠

Ω
---m

(A.24)

Figure A.8 shows the measured and MATLAB simulated values of the frequency
response of the near-end crosstalk of the circuit presented. As it can be observed
both results are very close.

Figure A.8: Measured and MATLAB simulation values of the near-end crosstalk.

300

A.3. MTL programs validation

A.3.2 Field incident
For this sections the voltage induced on two conductors above an infinite groundplane (2 cm and 5 cm), when are illuminated by an uniform external plane wave of
1 V/m, traveling parallel to the ground plane is studied. This is shown in figure A.9.

Figure A.9: Circuit under study.
The terminal configurations, characterized as a generalized Thevenin equivalent
structures are:Vˆ s = 0
1(A.25)Vˆ L = 00(A.26)Zˆ L = 500 0 Ω
0 1000(A.27)Zˆ S = 100 0 Ω
0 500(A.28)For the presented example, it has been considered a lossless line of 1 meter and theper unit length parameters for this configuration are:301

Appendix AµH
L = 0.97538 0.0955 -------0.09555 1.0119 m(A.29)

pF
C = 11.507 – 1.09 ------– 1.09 11.08 m(A.30)

Figure A.10 shows the measured and MATLAB simulated values of the frequency
response of the near-end crosstalk of the circuit presented. As it can be observed
both results are very close.

Figure A.10: Measured and MATLAB simulation values of the near-end crosstalk.

A.4 References[1] C.R. Paul, "Analysis of multi-conductor transmission lines", 1992, ISBN-0-471-02080- X.
[2] C.R. Paul, "Introduction to Electromagnetic Compatibility", NY: Wiley-Interscience,
1992, ISBN-0-471-54927-4.

302

Appendix B - Transients
B.1 Dynamic load impedances of Tracker FEE
The dynamic impedance of the APV25 during the transient analysis is calculated
based on some measurements performed in Aachen by Waclaw Karpinski. These
measured values are shown in figure B.1 and figure B.2. These plots show the U-I
curves of a group of 4 APV25 -TOB hybrid.
The first plot shows the FE-hybrid dynamic impedance on the V250-line varies fromabout 3 to 7 ohms.

Figure B.1: TOB FE-Hybrid impedance - 2.5V line
Figure B.2 shows the FE-hybrid dynamic impedance on the V125-line. It varies fromabout 2.5 ohms to 8 ohms below 1.15 V. However, it behaves like a current source
above 1.18 V and has an infinite impedance value.

303

Appendix B

Figure B.2: TOB FE-Hybrid impedance - 1.25V line

B.2 Transient lumped model validation
To validate the lumped model of the cable used in the transients simulations
analysis, Pspice simulations using 50 meters of the tracker power cable (MSC) havebeen performed using the transient lumped model. These results have been
compared with real measurement performed by Robert Hammarstrom.
The configuration of the test is shown in figure B.3. The HV and sensor wires are left
open during the test. The 2.5V line is loaded with a dynamic load which is set to 10
A at 2.5 V, with a damping resistance of 2.5 ohms and a capacitor of 125 µ f . This
damping impedance is calculated based on the dynamic impedance ofHybrid-4APV25 shown in figure B.1. At load level, a dynamic current step isinjected into the V2.5V line, which generates a step of 1 A in this line. The load
power is supplied by a voltage source at 5 V.

V1 .25 V
C 1 . 25 V

I 1 .25 V
I Load = 10 A − 11 A

I 2 .5V

C ableV 2 . 50 V
C 2 .5 V

R 2 .5 V

Figure B.3: Circuit used in these test.

304

B.2. Transient lumped model validation

For the test, the V1.25 line is left open in both sides of the cable and two steps of 1 A
are applied to the line, one positive and the other negative. The voltage and thecurrent on the V2.5 line are measured and compared with the Pspice simulation
based on lumped transient model1.
Figure B.4 and B.5 shows the simulated and measured values for the test. The
voltage at load level is measured with channel 2 ( V 1 -200 mV/div), the current in
the V2.5 is measured with channel 4 ( I 2.5 V -0.5 A/div) and the current step isµs
measured with channel 1 (1 A/div). The time scale is 200 -------- .
div
400mV

V-2.5V

200mV-0mV-200mV-400mV-600mV
V(R5:2)-2.5
0.5A

0A

I-2.5V

-0.5A

-1.0A
SEL>>
-1.5A
0s

0.2ms
I(L1A)+11

0.4ms

0.6ms

0.8ms

1.0ms

1.2ms

1.4ms

1.6ms

1.8ms

2.0ms

Time

Figure B.4: Simulated voltage and current in the 2.5V line

Figure B.5: Measured voltage and current in the 2.5V line
1 See section 7.3.2

305

Appendix B

For the second test both sides of the 1.25V line are connected through a capacitor of
C 1.25 V = 66 µ f . Again, the V2.5V line is loaded with a dynamic load which is set to
10 A at 2.5V with a damping resistance of 2.5 ohms and a capacitor of 60 µ f . The
voltage and the current in the 1.25V line as well as the voltage on the 2.5V line havebeen measured.

-0 m V-2 0 0 m V-4 0 0 m V

V-2.5V

-6 0 0 m V-8 0 0 m V
V (R 5 :2 )-2 .5
100m A0A

I-1.25VSEL>>-1 0 0 m A
I(L 2 A )100m V

V-1.25V0V-1 0 0 m V
0s50us
V (R 6 :1 )-V ( R 4 :2 )

100us

150us

200us

250us

300us

350us

400us

450us

500us

T im eFigure B.6: Voltage and current at the 2.5V & 1.25V lines - Simulated values
Figure B.7 shows the measured values of the test. The voltage at load level ismeasured with channel 2 ( V2.5 V -200 mV/div), the current I 1.25 V is measured withchannel 4( I 1.25 V - 0.1 A/div), the voltage at V 1.25 V line is measured with channel 3( V 1.25 V -100 mV/div) the current step is measured with channel 1 (1 A/div). Theµstime scale is 50 -------- .
div

Figure B.7: Voltage and current at the 2.5V & 1.25V lines - Measured values306

B.3. ATO fuse data sheet

B.3 ATO fuse data sheet

Blade Terminal & Special Purpose Fuses
Low Voltage

ATO® Fuse Fast-Acting Type

U® L

®

Designed and originated by Littelfuse for the automotive industry, theATO fuse has become the original equipment circuit protection standard
for foreign and domestic automobiles and trucks. Readily identifiable andeasily replaced, this fuse can be specified for a variety of low voltage
electronic applications.

ELECTRICAL CHARACTERISTICS:% of Ampere
Rating
110%
135%
200%
350%

AmpereRating
1– 40 Amp
1–2 Amp3 – 40 Amp
1–2 Amp3 – 40 Amp
1–2 Amp3 – 40 Amp

OpeningTime
100 Hours Minimum.50 sec., Min.; 600 sec., Max.
.75 sec., Min.; 600 sec., Max.
.10 sec., Min.; 5 sec., Max.
.15 sec., Min.; 5 sec., Max.
.020 sec., Min.; 0.5 sec., Max.
.080 sec., Min.; 0.5 sec., Max.

Reference Dimensions
19.05(.75")

5.08(.20")

AGENCY APPROVALS: Listed by Underwriters Laboratories (1– 40
amperes). Certified by CSA (3 – 30 amperes).

12.19(.48")DESIGN STANDARDS: U.L. Standard for Automotive Blade Type
Fuses. SAE (Society of Automotive Engineers) J1284.

6.35(.25")PATENTEDCOLOR-CODING: Autofuse® fuses are color-coded for easyamperage identification.

4.064(.16")
14.48(.57")

INTERRUPTING RATING:
1000A @ 32VDC

0.66(.026")ORDERING INFORMATION:
Voltage
Body
Rating
Color
(VDC)
Code
32
Black
32
Grey
32
Violet
32
Pink
32
Tan
32
Brown
32
Red
32
Blue
32
Yellow
32
Natural32
Green32
Blue Green
32
OrangeNominal Cold
Resistance
Ohms
0.123
0.050
0.031
0.023
0.018
0.011
0.00770.00480.00330.00250.00190.00160.0014

Minimum
Melting I 2 t(A2 Sec.)
.4
1.4
7.4
14
26
60
115
340
520
1080
1510
2280
3310

Average Time Current Curves
3A
4A
5A
7 1/2A
10A
15A
20A
25A
30A
35A
40A

AmpereRating
(A)
1
2
3
4
5
71/2
10
15
20
25
30
35
40

1000

100
TIME IN SECONDS

Catalog
Number257 001
257 002257 003257 004
257 005
257 07.5
257 010257 015257 020257 025257 030
257 035
257 040

10

1

0.1

0.01
1

10
100
CURRENT IN AMPERES

1000

Reference pg. 344, ATO® Fuse Clip for P.C. Board mounting.

318

w w w. l i t t e l f u s e . c o m

Figure B.8: ATO fuse data sheet

307

Appendix B

B.4 TVS validation
A normalized pulse1 of 340 V has been applied with a surge generator (UCS-500) on
the input filter2 of the real HCAL FEE to verify the validity of the parameters shownin the model. Figure B.9 shows the measurement voltage on the filter and figure
B.10 shows the simulated voltage on the filter.

Figure B.9: Voltage at the input FEE filter - Measured values

9.0V

8.0V

7.0V

6.0V

5.0V

4.0V

3.0V
0s

50us

100us

150us

200us

V(Lta:2)
Time

Figure B.10: Voltage at the input FEE filter - Simulated values

1 See figure 7.18 and section 7.3.3.1
2 See figure 7.20, section 7.3.2.2.1 and 7.3.3.2 for TVS and filter details
308Appendix C - Current probes

Figure C.1: Transfer impedance of current probe - EMCO-91550-1.

Figure C.2: IL of bulk injection current probe - Fischer F-120-9A.

309

Appendix C

310

Glossary• AC - Alternative Current• ADC - Analogue Digital Converter• ALICE - A Large Ion Collider Experiment• ATLAS - A Toroidal LHC ApparatuS• CCU - Communication and Control Unit
• CERN - European Center for Nuclear Research• CI - Common Impedance• CM - Common Mode
• CMS - Compact Muon Solenoid• CSC - Catode Strip Chambers• DC - Direct Current• DM - Differential Mode
• DT - Drift Tubes• ECAL - Electromagnetic Calorimeter• EMI - ElectroMagnetic Interference• EMC - ElectroMagnetic Compatibility• ENC - Equivalent Noise Charge• fC- Femto-coulomb• FE - Front End• FEE - Front End Electronics• Gnd - Ground• HCAL - Hadron Calorimeter• HEP - High Energy Physics• HF - High Frequency• HPD - Hybrid Photodiodes• HV - High Voltage• IL - Insertion Loss
• LF - Low Frequency• LHC - Large Hadron Collider

311

Glossary• LHCb - Large Hadron Collider beauty• LIC- Low Impedance Cable• LSB-Less Significant Bit• LISN - Line Impedance Stabilization Network• LV - Low Voltage• MSC - Multi-Service Cable• MTL - Multi-conductor Transmision Line
• RPC - Resistive Paralell Chambers• PACE - Pre-shower Analog Cms Electronics• PE - Protective Earth cable• PLL - Phase Lock Loop
• PMT - Photo-Multipliers• PS - Power Supply• QIE - Charge Integrator and Encoding chip
• RBX - Read Out Box• RF - Radio Frequency• RMS - Root Mean Square• SMPS - Switch Mode Power Supply• STP - Shieled Twisted Pair
• TeV - Teraelectron volt (energy)• TF - Transfer Function• THD - Total Harmonic Distorsion• TOB - Tracker Outer Barrel• TVS - Transient Voltage Suppresor• VME - Versa Module Europe.
• ZCS - Zero Current Switch• ZVS - Zero Voltage Switch

312