Analog Design Journal

Signal Chain

Precision signal-conditioning solutions formotor-control position feedback
By Ralph Oberhuber
Precision Amplifiers

Introduction

year (about 10 million coming from encoders), reflecting
the majority of implementations in the industry.
In the encoder world, analog signal processing is still thedominant approach for decoding incremental position
information using sinusoidal signals with a peak-to-peakvoltage of 1 VPP. This approach requires the lowest
amounts of both software implementation effort and cost.
For absolute position encoding, serial protocols have beenon the rise, but require more effort (and thus cost) interms of hardware and software. A hybrid implementationof digital and analog is still common. In the resolver world,
analog frequency-modulated signals can decode angle andposition in harsh environments, which makes them ideal
for factories or automotive applications.

Electric motors are an ever-increasing part of our lives,
not only in automated factories and electric vehicles, but
inside homes as well—in living rooms and even our children’s favorite toys. Picture an automotive factory where
robots pick up heavy parts or engines and place themexactly into car assemblies, requiring precise motor control
as the robots employ variable amounts of speed and torque.
These types of applications are driving advances in electronic technology that enable precision control duringreal-time operations. As intelligent motor drivers, high-end
power supplies and signal processing work together todeliver precise speed and torque,[1] ­position-sensing feedback is a key subsystem.
More complex and precise motor-control systems require
more advanced position-feedback implementations. For
example, a modern standard-robot configuration can besix motors per axis, with resolver feedback required foreach motor.

Position-feedback system requirements
Figure 1 shows a typical analog position-feedback architecture. Either a sine/cosine encoder or a resolver’s
secondary windings can generate the 1-VPP differential
input signals (A+, A–, B+ and B–), with the main difference being the signal’s frequency range of interest (0 to400 kHz for optical encoders versus 10 kHz up to 20 kHz
for a typical resolver), as well as the amplitude. The idea
is to level-shift, amplify and filter the signals of interest
while rejecting unwanted common-mode disturbances andconverting the differential inputs into single-ended

Position-sensing technologies
Although magneto-resistive and hall-sensing technologies
are starting to get traction in automotive and integrated
motor applications, the focus of this article is on angle/
position encoders (incremental and absolute) and resolvers.
Together, there are as many as 13-million channels per

A+
A–

Sin

A

B+

Connector

B–

Resolveror
Sin/Cos
Encoder

Cos B

C+Marker,C– or
Excitation
Feedback

C

AOUT

BOUT

COUT

AIN

ADC-A Digital Out

AIN

ADC-B Digital Out

AIN

ADC-C Digital Out

Host Processor Connector

Figure 1. Typical motor-control system with position feedback

MCU

+ EXC
2

PWM

Resolvers
Only

– EXC
2x Excitation Amp

Texas Instruments

Active Low-Pass1

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Signal Chain
Table 1. System-level requirements for a position-feedback
signal path

outputs (AOUT and BOUT), all simultaneously. After conditioning the input signals, the resulting signals can bepresented to an analog-to-digital converter (ADC) for
accurate digitization and extraction of the position/angle
information. References 2, 6 and 7 offer a detailed explanation on how the angle/position of the motor is extracted
from the difference between the instantaneous voltage
readings on channels A and B. A third channel, C, isfrequently used as either a marker (encoder case) or excitation feedback path (resolver case).
Table 1 is an example set of system-level requirements.
The specifications given in the table reflect the minimum
performance needed in the signal chain to guarantee theaccuracy of an angle measurement with the targetedmaximum phase error of <0.2%.[2] While this analysis
focuses on an angular measurement, the procedure iseasily applicable to a linear position measurement as well.
The mismatch of both offset and gain between the twochannels (sine/cosine) contributes significantly to theoverall phase error due to imbalance in the signal­propagation.[3]Drift over temperature is a critical concern as wellbecause calibration at more than one temperature point istypically prohibited due to cost reasons. The commonmode requirement guarantees that disturbances up to±10 V are suppressed to a level below 0.1% of the fullscale-range (FSR) input signal, which corresponds to a
maximum tolerable rejection ratio of 0.1% × 1 VPP/10 VPP =100 µV/V, or 80 dB. The common-mode disturbance could
be created due to shifts in the signal ground level fromlong cables between the sensor and signal-conditioning
circuit’s printed-circuit board. Another disturbance source
could be due to coupling from power-carrying wiringlocated close to the signal cables from the encoder to themotor shaft. There might be additional requirements suchas noise and slew rate, however, those are omitted forsake of simplicity.

System Parameter

Target Value

Impact on
Phase Error

Ambient temperature–40°C to 125°C—

Signal frequency

0 to 500 kHz (encoder)
10 kHz ±1 kHz (resolver)—

Input signal
amplitude/offset

1 VPP at 2.5-V offset or
0‑V offset (encoder),4 to 7 VRMS (resolver)—

Offset error of A and B(across temperature)

<0.1%

0.05%
(0.18°)

Gain error of A and B(across temperature)

<0.1%

0.05%
(0.18°)

Phase shift between A
and B

90° + 0.36’’ (<0.1%)

0.1%(0.36°)

Common-mode rejection(assuming up to ±10-V
perturbation)>80 dB

0.1%(0.36°)

Total root-mean-squarephase error

—

0.16%
(0.57°)

Signal-chain device requirements: DC
The first step is to break down the system-level requirements into detailed device specifications for the individual
analog signal-path components. For the signal-path implementation shown in Figure 2, input buffers A1 and A2 are
optional, and are frequently omitted in case the signal
source has low output impedance. In applications requiring high input impedance, an instrumentation amplifier
such as the INA821 or INA819 from Texas Instruments
might be a better choice (also see Table 3 on next page).
In a practical implementation, additional filtering and/or
voltage limitation/clamping could be added to the circuit
shown in Figure 2.[2] Scaling the input signal amplitude of

Figure 2. Precision signal-path implementation
VCC

Buffers (optional)

+

A–

–

OPA192 +4 Resistorsor
INA592
INA137

V+

–IN

SENSE

A1
R1

R2
–
+

+

A+

–

A2

A3

+IN

VOUT

AIN

ADC –A Digital Out

REF
R4 = R2R3 = R1V–

VOUT =(For simplicity, only the A path is shown.)

Texas Instruments

2

R2
× ( VA + − VA - )R1

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Analog Design Journal

Signal Chain
Table 2. Device-level requirements for difference-amplifier stage
with position feedback

1 to 2 VPP into a full-scale input signal that maximizes the
ADC dynamic range typically requires a gain in the rangeof 1 to 4 V/V, depending on the ADC chosen. In the
resolver case, depending on the loss factor from theprimary to the secondary coil and the resulting output
amplitude, it is sometimes necessary to scale down the
4- to 7-VRMS input signal to accommodate the ADC input
range.
In order to present the differential input signal to the
ADC in a single-ended fashion, amplifier A3 is connected
in a well-known “difference-amplifier” configuration that
could be either integrated into one integrated circuit or
implemented with a stand-alone operational amplifier anddiscrete, well-matched resistors. In both cases, the gain isset by the resistor ratios and the offset- and gain-error
requirements translate directly into an offset and gain
specification for difference amplifier A3 in combination
with resistors R1, R2, R3 and R4 (shown in Table 2).
Equation 1 translates the percentage offset error into an
absolute offset-error voltage based on the 1-VPP FSR:

Meets
Spec.?

Ambient temperature–40°C to 125°C–40°C to 125°C

Yes

Gain

Typically in the
0.5 to 4 range

0.5 or 2

Yes

Gain error (A3, B3)

<0.1%

0.03%

Yes

Offset error (A3, B3)<1 mV

40 µV max

Yes

Offset drift (A3, B3)<3 µV/°C

1.5 µV/°C max

Yes

Bandwidth (3 dB)>500 kHz

2.5 MHzYes>80 dB>80 dB acrosstemperaturesYes 1 + R 2 / R1
CMRR (dB) = 20 log  4T / 100 (3)

where T is the resistor tolerance in percent (%).
As shown in Reference 4, even for an external resistor
tolerance of 0.01%, the CMRR performance will be limited
to 74 dB due to resistor mismatch. This would make animplementation with discrete resistors achieving the target
specification of >80-dB CMRR extremely costly. At thetime of this writing, an array containing two matched
10-kΩ resistors with a tolerance of 0.01% is (US) $11.55.
On the other side, a modern monolithic difference
amplifier such as the INA592 provides a CMRR of >80 dB
across all process variations and temperatures by using
thin-film resistor technologies, which enable on-chip resistor matching at the 0.001% level.

(2)

As an alternative to the INA592, a discrete implementation of the difference-amplifier stage could be tried. Thismeans using a stand-alone operational amplifier such as
the OPA192 in combination with fully discrete resistors, orTexas Instruments

INA592even a matched resistor array. The common-mode rejection ratio (CMRR) is dominated by a mismatch toleranceof resistors R1, R2, R3 and R4 (assume R3 = R1 andR4 = R2). Equation 3 gives the CMRR, according toReference 4.

1V(1)= 1 mV
1000
Equation 2 obtains the temperature drift specification
for A3, also shown in Table 2. It becomes clear from thefarthest right column in Table 2 that the INA592 difference amplifier from Texas Instruments is an excellent
choice for this solution, as it meets or exceeds all specifications needed.
1 mV × 0.5= 3 µV / °C
125 − ( −40) °C

Maximum
Specification

Common-mode­rejection ratio (CMRR)

VOS _ max ( A, B ) = 0.1% ×1 V =DriftOS _ max <Device parameter3

ADJ 3Q 2019

Analog Design Journal

Figure 3. Spice simulation circuit and results
INA592

Mismatched resistor:
R2 + ∆R

R1 6k+

Common Mode
Perturbation:
400 mVpp@ 5 kHzVG1

R2 12.1k

VIN1

C2 6p

VCC

+

VEE
Differential
Signal
1 Vpp@ 50 HzVG2

VEE C3 6f

+

VIN2

VEE

Vout
+

U1 OPA192

VCC
R3 6k

V1 15

R4 12kV2 15(a) TINA-TI™ simulation schematic

1.01
Vout
= G × (VIN2 – VIN1)= 2 × (VIN2 – VIN1)

Voltage (V)To illustrate the performance differences between the discrete and themonolithic difference-amplifier solutions,Figure 3a is a schematic of TINA-TI™
Spice simulations. The circuit is essentially the same as that shown in Figure 2,
but without a buffer, and the resistors
are configured to achieve a gain of 2. The
resistors are intentionally mismatchedby modifying the value of R2, which
results in various levels of mismatch
ranging from 10% to 0.01%. In a practical application, a 1% or 0.1% mismatchcorresponds to a discrete implementation versus a mismatch of only 0.001%,
as in a modern monolithic difference
amplifier like the INA592. The input
signal is a differential signal of 1 VPP at
50 Hz, which “rides” on a common-mode
perturbation of 400 mVPP at a 5-kHz
frequency. Figure 3b shows a full-scale
plot of the input signal VIN2 and theoutput voltage, while Figure 3c shows azoomed-in version of the output voltage
for four different simulations withvarying mismatch. Clearly, the commonmode perturbation is significantly
suppressed for the cases with tighter
matching of the resistors.

Signal Chain0

VIN2= 400 mVPP at 5 kHz+ 1 VPP at 50 Hz
–1.01
0

10

20

30

40

50

Time (ms)(b) Input/Output simulation results
1.07

Output Voltage (V)

10% R2 Mismatch
1% R2 Mismatch

0.1% R2 Mismatch

972.60m

0.001% R2 Mismatch
871.97m
31

32.5

34

35.5
Time (ms)

37

38.5

40

(c) Zoomed-in simulation results

Texas Instruments

4

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Analog Design Journal

Signal Chain

Figure 4. AC coupling mechanism in motor systems that
results in high-frequency common-mode perturbations

3-phase PWM switchedvoltage in power cable:
e.g. 10 kV at 8/16/32 kHz
Common-mode noiseon shield. Can be up to±20 V, pulse width < 1 µs

Servo Drive
Power Module

Gear

Shielded encoder
cable, up to 100 m
and more
Angle
SensorAC
Motor

Position Feedback
ModuleResolveror Sin/Cos
Encoder

Differential sin/cos
signals in encoder cable.
Resolver: 4 to 7 VRMS
Sin/Cos encoder: 1 VPP

Signal-chain device requirements: AC

Figure 5. AC common-mode rejection:
the INA592 vs. the INA137

One important aspect that could be easily overlooked in
the design of the precision signal path is the immunity
against high-frequency perturbations.
High-frequency impulse, common-mode noise is typically coupled from the motor wires that carry switchedpulse-width modulation signals into the resolver cables, ordirectly into the motor housing; see Figure 4.[5] To connect
the wires to the motor, such as inside a robot, the shield of
the motor cable might be open for up to 20 cm. The
common-mode disturbance therefore is typically AC;
pulses are ≈1-µs wide and have an amplitude of ±20 V. In
this case, the common-mode rejection at higher frequencies between ≈400 kHz up to 1 MHz is critical.
Figure 5 compares the CMRR-versus-frequency performance of the INA592 (small-signal bandwidth = 2.5 MHz)and INA137 (small-signal bandwidth = 4 MHz). The
INA137, which is based on the higher-bandwidth amplifiercore, has an improved CMRR in the 400-kHz to 1-MHz
signal range and therefore will perform better in suppressing unwanted high-frequency impulses, but at the cost of aworse DC precision from offset and offset drift; see Table 3
on the next page.

Texas Instruments

SignalAnalog Processing

120

Rejection Ratio (dB)

100

INA592, G = 1/2
INA137, G = 1/2

80
60
40
20
10

5

100

1k
10 k
Frequency (Hz)

100 k

1M

ADJ 3Q 2019

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Signal Chain

Table 3. Solution performance comparison using different component choices

Parameter

OPA192 Plus Four Resistors(matched 0.01%)

INA592

INA137

Output-referred offset VOS (max)

80 µV

80 µV

2 mV

CMRR at 1 kHz<74 dB>80 dB>74 dB

CMRR at 500 kHz<50 dB<50 dB

70 dB

Board size

Large

Small

Small

>$5/channel$1.30/channel$1.27/channel

Price (1,000-unit quantities)

Other
Recomended
Devices
THS4551
THS3531A
OPA365
OPA322
PGA411
INA821
INA819Conclusion

References

Table 3 compares three different options for implementing
the precision signal path: discrete (the OPA192 plus four
resistors) and two alternatives using the INA592 and
INA137 difference amplifiers. Table 3 summarizes thetrade-offs discussed in this article. As outlined earlier, theINA592 is the best option whenever offset, offset drift and
CMRR are important. For those cases where CMRR at
500 kHz and beyond is the major concern, consider the
INA137. There are other options like using higher bandwidth amplifiers if an extremely fast response is required(see other device recommendations in Table 3), or evenusing a fully-differential amplifier when interfacing withdifferential-input ADCs.
In the analysis for this article, the focus was solely on
the example of a differential-to-single-ended signal chain.
By starting from the system-level requirements, a systematic approach has been presented on how to evaluate thecomponent-level requirements, while considering tradeoffs in terms of performance, size and cost.

1. Harald Parzhuber, “High precision in motor drive­control enables industrial advances,” TI white paper(SLYY117), June 2017.
2. V. Pizzolante, M. Staebler and P. Stauder, “Interface to
Sin/Cos Encoders With High-Resolution Position
Interpolation,” TI Designs (TIDUA05A), July 2015.
3. Jiri Panacek, “Discrete Resolver Front-End Reference
Design With C2000™ Microcontroller and ±0.1°
Accuracy,” TI Designs (TIDUDP0), March 2018.
4. John Caldwell, “Designing for high common-mode­rejection in balanced audio inputs,” TI Analog Design
Journal (SLYT737), 1Q 2018.
5. Martin Staebler and Josef Mieslinger: Personal­communications.
6. K. Mogensen and M. Staebler, “Interface to a
HIPERFACE Position Encoder,” TI Designs(TIDUB64A), March 2016.
7. K. Mogensen V. Pizzolante and M. Staebler, “Interface toa Sin/Cos Encoder with Sitara AM437x,” TI Designs(TIDUA77A), September 2015.

Related Web sites
TINA-TI™ SPICE-based analog simulation program
Product information:
INA821, INA819, INA592, INA137, OPA192
THS4551, THA3531A, OPA365, OPA322, PGA411

Texas Instruments

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Analog Design Journal

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