AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATIONHIGH VOLTAGE BUCK-BOOST LED HEADLAMP DRIVER
USING ALLEGRO A6271-1
By Shashank Wekhande, Vashist Bist, and Sarang Gadgil
Allegro MicroSystems

ABSTRACTINTRODUCTION
The A6271-1 is an automotive AEC-Q100 qualified DC-DC
controller that provides a programmable constant-current
output for driving high-power LEDs in series. The controller isbased on a programmable fixed-frequency, peak current-mode
control architecture. The A6271-1 offers an internal switchingfrequency dither option that helps to reduce electromagnetic
interference (EMI). The DC-DC converter can be configured invarious switching configurations including boost, buck-boost(VIN referred boost), buck (ground referred switch), and SEPIC
configuration.
The A6271-1 is rated for a maximum output voltage of 53 V
which limits the operation in buck-boost mode to three seriesconnected white LEDs while maintaining 40 V load dump
support. While a proper selection of the external MOSFET canallow for high output voltage, the output voltage is limited by
the voltage rating on the LP and LN pin. This limitation can beaddressed by implementing a buck-boost converter using theA6271-1 as shown in Figure 1. Theoretically, this configuration
can be used for any output voltage by proper voltage rating
selection of the MOSFET (SW), Diode (DSW), and output
capacitor (COUT) but output voltage (VOUT) is ultimately limitedby the maximum duty cycle of the switching converter.

RESD

VIN

LN
VREG

CREG

-

RDR

CPWM

SP
OVUV

LP

LB/HB
Control

RC1

RSG
RSP

COUT

CSLW
RLED

DOV1
ROV1

CC2 CC1
SG

PWMIN

RLB1

VLED

DZLB
RLB3 SWLB

COMP

+
-

RLB2

DZOV

A6271-1

CIREF

ILED

CBYP

OSC

IREF
RIREF2DSW
7 LEDs

DITH

FAULT
DR

LSW
CIN

LP VINGND

RFLTRIREF1

+

CSEN

ROSC

LP

RDITH

The proliferation of light-emitting diode (LED) drivers in various
automotive applications over the past decade has been driven
by a need for efficient lighting and aesthetics. Multiple seriesconnected LEDs are used for applications such as daytime runninglights (DRL) and headlamps with different LED configurations
and different driving currents. In such applications, buck-boost
topology is commonly used to drive LEDs over a wide range of
input and output voltages. This application note discusses thefeatures and application of the Allegro A6271-1 switching LED
controller in automotive high-power applications when drivinga high number of LEDs. A detailed design procedure for theA6271-1 in this application as well as details about the behaviorof the device during steady-state operation, transient analysis,
and behavior of various protection features are also discussed.

VIN
VBAT

VIN
ROV2
OVUV
ROV3

SW

RSEN

ROV4

SWOVP

OVUV

Figure 1: High-voltage buck-boost configuration using A6271-1.

In Figure 1, the inductor (LSW), power MOSFET (SW), andasynchronous diode (DSW) form the part of a buck-boost powerstage that provides output voltage across the capacitor (COUT).
High-beam or low-beam operation can be achieved by either
turning on all LEDs (10 LEDs) in a string or only a few LEDs
(7 LEDs) and short-circuiting the rest. An external P-channel
bypass MOSFET (SWLB) connected across the portion of an LED
string is used for shorting the LEDs during low-beam operation.
Due to the limited response time of the power stage controller,
LEDs may experience a large deviation in the current duringhigh/low-beam transitions. It is important to minimize these
LED current deviations to eliminate any undesirable flicker.
LEDs current deviations during high/low-beam transitions canbe minimized by slowing the turn-on and turn-off rate of the
bypass MOSFET (SWLB). The gate-drain capacitor (CSLW) acrossthe bypass MOSFET (SWLB) is added to limit the gate slew rateduring the high-beam to low-beam transition to limit the inrush
current. The components such as the gate protection Zener(DZLB) and resistor (RLB2) are used for the gate protection of the
P-channel bypass MOSFET (SWLB).

AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

COUT

VIN

VOUT

VLED
LSW
VIN

DSW

+
-

SW

Figure 2 (a) shows the simplified power stage of the buck-boost
converter and Figure 2 (b) and Figure 2 (c) show the operationof the buck-boost converter in continuous conduction mode
(CCM) during switch-on and switch-off period respectively.
Figure 3 shows the waveform of the buck-boost converter in
CCM mode over a complete switching cycle.

VOUT

VSG
(a)

ΔiSW
iSW

COUT

VIN

VOUT

VLED

iLSW

LSW
VIN

DSW

+
-

SW

VOUT

VOUT

(b)

VOUT

VLED
LSW
VIN

DSW

+
-

∆VDC
tON

tOFF

1/fSW

Figure 3: Waveforms of the operation of buck-boost converter in CCM.

COUT

VIN

ΔiLSW

SW

VOUT

(c)

Figure 2: Buck-boost converter (a) Simplified power stage, (b) Operation during the switch-on period, (c) Operation during the switch-off period.

OPERATION
The A6271-1 is a constant frequency, peak current-mode
controller. The switching frequency is set with a resistor on the
OSC pin, and an optional dither feature can be enabled byconnecting resistor to the DITH pin for better EMI performance.

The details on the operation of the buck-boost converter arepresented below.
Mode-1 (Switch Turn-on): As shown in Figure 2 (b), when theswitch (SW) turns on (tON), the energy is stored in the inductor(LSW) and capacitor (CSW) provides the required energy to theLEDs. During this period, the current through the switch issensed by the sense resistor (RSEN as shown in Figure 1). Whenthe sense current becomes higher than the internal COMP
reference, the controller turns off the switch.
Mode-2 (Switch Turn-off): As shown in Figure 2 (c), during theswitch turn-off period (tOFF), the diode (DSW) conducts, andthe inductor provides the required energy for charging output
capacitor (COUT) and LEDs. Energy stored in the inductor (LSW)
is released and therefore the inductor current reduces in thismode of operation as shown in Figure 3. The output capacitor
charges, and the voltage of the output capacitor (VOUT) increasein this mode of operation.
Considering the input voltage as VIN, output voltage as VOUT,
and the switch turn-on and turn-off time as tON and tOFF, bybalancing the volt-second across inductor LSW in CCM mode is:955 PERIMETER ROAD • MANCHESTER, NH 03103 • USA+1-603-626-2300 • FAX: +1-603-641-5336 • ALLEGROMICRO.COM

2

AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

Equation 1:

where ROSC is in kilo-ohms (kΩ) and fSW is in kilohertz (kHz).
VIN × tON = (VOUT – VIN) × tOFF

Replacing off time, tOFF as (t – tON) and duty (D) as tON/t as,
Equation 2: 1 
VOUT =  VIN
1− D This is a typical boost converter equation showing how output
voltage (VOUT, with reference to ground) is linked with the input
voltage (VIN).

2. Select the appropriate switching frequency dither to assistin minimizing EMI emissions. In this configuration, the main
oscillator can be dithered so that the energy is spread overa defined frequency band. The defined frequency band iseffectively the minimum and maximum switching frequency
selected. This frequency is varied above and below theselected oscillator frequency and is set via a resistor RDTH.
The frequency band can be selected as follows:
Equation 7:
R ∆f SW = ± 22 ×  OSC  RDTH 

However, if the voltage across LEDs (VLED) is considered, whichis the difference of the output voltage (VOUT) and input voltage(VIN), the volt-second balance equation can be written as:

where ΔfSW is a plus/minus percentage change to the oscillator
frequency fSW.

Equation 3:
VIN × tON = VLED × tOFF

Replacing off time, tOFF as (t – tON) and duty (D) as tON/t in
Equation 3 as:
Equation 4:
 D 
VLED =  VIN
1− D 3. Set the LED current (ILED) by selecting voltage (VIREF) on the
IREF pin and LED sense resistor (RLED). For this application,
the IREF pin is directly connected to the VREG such that theinternal current limit reference is set to the 200 mV reference
level (VIDL). Therefore, the LED current (ILED) is selected as:
Equation 8:
I LED =

This shows that the voltage across the LED string (VLED) varies
with the input voltage in the buck-boost configuration.

where ILED is in amperes (A) and RLED is in ohms (Ω).

The LED current is sensed through LP and LN pins by sensing thevoltage across the current sense resistor (RLED). The LED current
is programmed by the LED sense resistor (RLED) as:

The current can also be set by using a potential divider(RIREF1 and RIREF2) and the voltage on the IREF pin (VIREF) canbe adjusted as shown in Figure 1. The brightness control of
LED in high-beam/low-beam operation can also be done byadjusting the IREF current (i.e., VIREF voltage) using an externaltransistor as shown in Figure 22.

Equation 5:
I LED =VIDL
RLED

where, VIDL is the current limit comparator reference voltage
with a typical value is ~200 mV. This level can be changed foranalog dimming by setting reference voltage on the IREF pin.

4. For a 100% duty, without any potential divider (RIREF1 andRIREF2), the soft-start time (tSS) is configured by selecting theappropriate value of capacitor on the IREF pin (CIREF) as:
Equation 9:

NOTE: It is advisable to insert a 150 Ω (RESD) resistor in series
with the LN pin, as shown in Figure 1, to protect the internal ESD
structures between LN and LP under certain fault conditions.

t × 1× 10−6
CIREF = SS
1.2

DESIGN PROCEDURE

If a potential divider (RIREF1 and RIREF2) is used on the IREF pin,then the IREF pin capacitor (CIREF) for the soft-start featureneeds to be configured as per the description presented in
the datasheet [1].

The design procedure for the A6271-1 is as follows:
1. Configure the device switching frequency (fSW) using thefrequency setting resistor (ROSC) as shown in Figure 1 as:
Equation 6:
f SW =

25690
ROSC

VIDL
0.2
=
RLED RLED

5. For this application, no external PWM control is used andthe device is configured the 100% duty with internal PWM
control. For achieving this a small 10 nF capacitor is placedon the PWMIN pin.

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AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

6. Select the output inductor (LSW) as:
Equation 10:

LSW =VIN ( min ) × D ( max )∆I LSW × f SW

where VIN (min) is the minimum input voltage, D (max) is the
maximum duty of the converter, ILSW is the peak-to-peakripple current in the inductor, and fSW is the PWM switchingfrequency. The inductor ripple current can be set to 20-40%
of the average current (IAVG) which is expressed as:

8. Select switch current sense resistor and components on the
COMP pin (control loop related) as per datasheet.
9. Select the appropriate setting for the overvoltage fault. Set
voltage on OVUV pin at nominal operating conditions toapproximately (VIN – 0.5 V). A simpler way to achieve thiscondition is to use a diode with forward voltage drop suchas 0.2 V < VF < 0.75 V as shown in Figure 1 (Diode DOV1).
Under normal operating conditions, the MOSFET (SWOVP)
is off, the voltage on the OVUV pin is set by diode DOV1, andresistor ROV1 provides a small biasing current for diode DOV1.
During an overvoltage event, the voltage on the OVUV pin
must be 1.1 V (typical) below the input voltage (VIN) to trigger
the overvoltage protection. In such a scenario, the Zener
diode DZOV conducts which turns on the MOSFET (SWOVP).
This pulls the voltage on OVUV pin below 1.1 V with respect
to VIN and triggers the overvoltage fault in the IC.

Equation 11:
I avg =

D ( max ) × I LED(1 − D ( max ) )Select appropriate saturation and RMS current specification
for inductor design.
7. The output capacitor (COUT) is designed as:

The values presented in the above design procedure can beselected using the standard A6271-1 design tool [2].

Equation 12:

COUT =I LED × D ( max )∆VLED × f SW

where ΔVLED is acceptable peak to peak LED voltage ripple.

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AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATIONEXPERIMENTAL SETUP DETAILSAn experimental board to drive 10 LEDs in high-beam and7 LEDs in low-beam for automotive headlamp application isshown in Figure 4. This application can be extended for similar
applications such as the tail/stop light and daylight running
lamp/position lamp. The circuit is configured in buck-boost
topology to operate 8 to 18 V input voltage range with 40 V
load dump. The associated schematic is shown in Figure 22
and the bill of material (BOM) is shown in Table 1.
A6271-1 Device

ILED
VOUT

VOUT = ~40 V

SW NodeFigure 5: Steady-state operating waveforms for 10 LEDs (high-beam) for an
input voltage of 12 V.
ILED
VOUT

VOUT = ~69 V

Input
Terminals

Inductor

Power MOSFET

Output Terminals

Figure 4: High-voltage experimental board using A6271-1.

STEADY-STATE OPERATION
Figure 5 and Figure 6 show the steady-state operation of the
buck-boost converter using the A6271-1 device in high-beam
operation at the input voltage of 12 V and 40 V respectively.
As shown in Figure 5, the output voltage is maintained at 40 V
for the input voltage of 12 V considering the voltage drop of
10 LEDs around ~28 V. Similarly, in Figure 6, the output voltageof the buck-boost converter is maintained at ~69 V during theload dump scenario input voltage of 40 V.
The low-beam operation of the buck-boost converter using
A6271-1 at the input voltage of 12 V and 40 V is shown in
Figure 7 and Figure 8 respectively. As shown in Figure 7, theoutput voltage is maintained at 32 V for the input voltage of
12 V considering the voltage drop of ~20 V for 7 LEDs. Similarly,
in Figure 8, the output voltage of the buck-boost converter ismaintained at ~60 V during the input voltage of 40 V.

SW NodeFigure 6: Steady-state operating waveforms for 10 LEDs (high-beam) for an
input voltage of 40 V.

ILED

VOUT

VOUT = ~32 V

SW NodeFigure 7: Steady-state operating waveforms for 7 LEDs (low-beam) for an
input voltage of 12 V.

ILED
VOUT

VOUT = ~60 V

SW NodeFigure 8: Steady-state operating waveforms for 7 LEDs (low-beam) for an
input voltage of 40 V.

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AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

85

High-Beam
Low-Beam

84

DeviceEfficiency (%)

83
82

MOSFET

81
80
79
78

Inductor

77
76

8

12

16

20
24
28
Input Voltage (V)

32

36

40

Figure 9: Plot of efficiency vs. input voltage for high-beam and low-beam
operation of buck-boost converter using A6271-1.
418

Figure 11: Thermal performance with ILED = 400 mA at 12 V input and room
temp for high-beam operation (10 LEDs).

High-Beam
Low-Beam

414

DeviceLED Current (A)410

MOSFET

406
402
398

Inductor

394
390

8

12

16

20
24
28
Input Voltage (V)

32

36

40

Figure 10: Plot of regulated LED current vs. input voltage for high-beam andlow-beam operation of buck-boost converter using A6271-1.

The efficiency and the LED current plot with respect to the input
voltage are shown in Figure 9 and Figure 10 respectively. During
the low input voltage operation, the input current becomes
high, which in turn increases the conduction losses of thesystem. Similarly, at the higher input voltage, the switchinglosses dominate due to the higher voltage stress on the device
which reduces the overall efficiency of the system.
Figure 11 and Figure 12 show the thermal images of the buckboost converter board operating at a rated input voltage of 12 V
driving 10 and 7 LEDs respectively. As shown in these figures,the maximum temperature on the board is around 61°C / 55°C
at room temperature with high-beam / low-beam operation at
a rated input voltage of 12 V.

TRANSIENT OPERATION
Figure 13 and Figure 14 show the transient behavior during thehigh-to-low beam and low-to-high beam respectively. As shownin Figure 13, as soon as the control input (yellow) is applied, theoutput voltage (pink) steps down to adjust for 7 LEDs (low-beam
operation). A limited inrush LED current (green) is observed withinthe ~12% overshoot limit. Similarly, during the low-beam to highbeam operation, an unnoticeable dip in LED current is observed
and the output voltage settles within a time duration of 2 ms.

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Figure 12: Thermal performance with ILED = 400 mA at 12 V input and room
temp for low-beam operation (7 LEDs).

Control In
VOUT
ILED

HB to LB Transition
Output Voltage Drop of ~10V

Current Overshoot ~11.5%

VCTH

Figure 13: Transient behavior during high-beam to low-beam transition.

Control In

VOUT

LB to HB Transition

Output Voltage Jump of ~10V

ILED
VCTH

Figure 14: Transient behavior during low-beam to high-beam transition.

6

AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATIONPROTECTIONSThe A6271-1 device integrates numerous protection schemes
and is described as follows. This application does not have output
undervoltage protection or overvoltage protection because ofmodifications required to operate the IC for higher output voltages.
However, with some additional circuitry, these two protection features
can be added with this high-voltage configuration as explained below.
OUTPUT OVERVOLTAGE PROTECTION (OPEN-LED): In an
unwanted scenario of an open LED, the output voltage rises as
the LED current feedback becomes zero. Therefore, the converter
operates in an open loop and increases output voltage in an
uncontrolled manner, and eventually rises to the breakdown
voltage of overvoltage protection Zener diode (DZOV) as shownin Figure 1. This Zener breakdown turns on the overvoltage
protection MOSFET (SWOVP) and the current starts flowing
through divider resistors (ROV2 and ROV3). This creates a voltage
drop and the overvoltage detection pin (OVUV) is pulled low
to trigger a fault inside the IC. During this scenario, the output
voltage is regulated to the OVP level. The converter returns tonormal operation when the fault is removed.
Figure 15 shows OVP operation with 10 LEDs operating at theworst-case scenario of input voltage as 40 V (load dump operation).
The overvoltage protection Zener diode (DZOV) is designed for~90 V breakdown voltage. The device operation is stopped as
the LED current (green) becomes zero as soon as OVP is detectedin an open LED scenario. Figure 16 shows the hiccup shutdown
period, as the overvoltage condition is applicable, and the drivertries to restart after the fixed hiccup period of 26.5 ms (tHIC).
OUTPUT UNDERVOLTAGE PROTECTION: Additional output
undervoltage protection is implemented as shown in Figure 17. In
this circuit, if output voltage (VOUT) goes below the Zener diodebreakdown voltage (DZUV), then the output transistor (SWUV) will
eventually turn off and turn on the SWOVP MOSFET. This in turngenerates a low signal on the OVUV pin of the A6271-1 device
and shuts down the operation.

VIN
VOVUV

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Output Voltage > 91V

ILED
Driver Operation Halt

Figure 15: Overvoltage detection during open-LED operation.

VIN
VOVUV Multiple OVUV
Detection
VOUT

Output Voltage > 91V

ILED

Driver Operation Halt

Figure 16: Hiccup operation during the overvoltage protection.

ILED

VREG

DSW

DZUV
RUV2
LEDs

DZOV

+
-

RUV3

RUV1
GateUV
SWUV

COUT
Output Undervoltage Detection
LP

INPUT UNDERVOLTAGE PROTECTION: If either input
undervoltage or the regulator (VREG) undervoltage scenario
occurs, then the low-side MOSFET drive (SG) is disabledand FAULTn is active. Both the input voltage (VINUV) and theVREG voltage (VREGUV) must rise above their respective turn-on
thresholds to enable the IC.
TSD (Thermal Shutdown): If the junction temperature of the
IC exceeds the overtemperature shutdown threshold, thelow-side MOSFET drive (SG) is immediately disabled, FAULTn
is active, and the IREF node (soft-start) capacitor is discharged
immediately. An auto-restart is performed under control ofthe soft-start capacitor once the temperature drops below theovertemperature minus the hysteresis level.

OVUV Detection

VOUT

RLED
VIN

DOV1
ROV1

ROV2
OVUV
ROV3

ROV4

SWOVP
GateUV

Figure 17: Additional circuit for the output undervoltage protection.

7

AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

LED SHORT: In the case of an LED short, the converter willcontinue to operate with the remaining LEDs and regulate LED
current. Figure 13 shows the transient behavior of the converter
during the transition from high-beam (10 LEDs) to low-beam(7 LEDs) which is the impact of shorting 3 LEDs.
OVERLOAD PROTECTION: In case of a LED overcurrent fault,
the high-side MOSFET drive (PWMOUT) and the low-side
MOSFET drive (SG) are disabled after two fault mask switchingcycles, FAULTn is active, the IREF node (soft-start) capacitor isdischarged, then the hiccup timer (tHIC) is initiated for a periodof 26.5 ms. After the hiccup period, an auto-restart is performed
under the control of the soft-start capacitor.

IMPACT OF SLEW RATE CONTROLThe slew rate control of the p-channel bypass MOSFET (SWLB)
is required to limit the inrush current in the LEDs during thehigh-beam to low-beam transition (turn-on of SWLB). Figure 18
shows the high-beam to low-beam transition without any slew
rate control on the P-channel bypass MOSFET (SWLB). As shownin this figure, a very high inrush current is observed, since theinductance on the load path (LEDs) is very minimal. The slew
rate of the gate voltage (yellow) and source voltage (blue) isvery fast which leads to a high inrush current and eventuallytrips the overcurrent fault.

VGate

The slew on the bypass MOSFET can be added either by additionof the gate-source capacitance or the gate-drain capacitance.
The addition of gate-source capacitance leads to a slower gate
voltage rise of the MOSFET due to the extra time required tocharge the additional capacitor. Figure 19 shows the waveforms
with added 1 µF capacitor between the gate-source of the bypass
MOSFET. As shown in this figure, the gate voltage (yellow) slew
becomes very slow, and a noticeable impact can be seen in
the slew of the source voltage. A current overshoot of ~23%
is observed for a high value of added gate-source capacitor.
A better approach of controlling the bypass MOSFET drainsource voltage slew rate is to control the operation in the Miller
region which can be done by adding the capacitance on thegate-drain pins of the bypass MOSFET. Figure 20 and Figure 21
show the impact of the addition of 100 nF and 47 nF capacitorson the gate-drain pin of the P-channel bypass MOSFET (SWLB).
The source pin slew rate (blue) is ~1000 µs (Figure 20) for a
100 nF gate-drain capacitor and ~500 µs (Figure 21) for a 47 nF
gate-drain capacitor. The current overshoot observed with a
100 nF gate-drain capacitor is around 7.25%, whereas a 14%
current overshoot is observed with a 47 nF capacitor on thegate-drain pin. As depicted here, the drain-source slew rate canbe directly controlled by changing the gate-drain capacitor andcurrent overshoot can be tuned according to the requirement.

VGate

Very Fast VGS Slew

VDrainVDrainVSource

Very Fast VDS Slew

VSource

ILEDHigh Current OvershootLeading to Driver Turn-off

ILED

Figure 18: High-beam to low-beam transition with no added capacitance tothe bypass MOSFET.
VGate

Slow VGS Slew

Slow VDS Slew

Current Overshoot ~7.25%

Figure 20: High-beam to low-beam transition with added gate-drain capacitance of 100 nF to the bypass MOSFET.
VGate

Fast VGS Slew

VDrainMedium VGS Slew

VDrainVSource

Fast VDS Slew

VSourceMedium VDS Slew

ILED

Current Overshoot ~23%

ILED

Current Overshoot ~14%

Figure 19: High-beam to low-beam transition with added gate-source capacitance of 1 µF to the bypass MOSFET.

Figure 21: High-beam to low-beam transition with added gate-drain capacitance of 47 nF to the bypass MOSFET.

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AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

ROV1

10 kΩ

US1D

ROV2

4.2 kΩ

OVUV

VIN

DIN2

1

DZOV

FB1

LIN

VAN
91 V Zener

2

"GND"

30 µH, 4 A
Common Mode Choke

TVS
50 V

CIN1

CIN2

CIN3

4.7 µF
50 V

100 nF
50 V

4.7 µF
50 V

CIN4

CIN5

4.7 µF
50 V

100 nF
50 V

Overvoltage Protection

Supply Side Filtering

X?A

VIN

VIN

VIN
VREG

LP

MBRS2100LP
CSEN
47 nF, 16 V

1 µF
16 V

LN
PWMOUT

FAULT

RESD
150 Ω

VIN

COUT1

COUT2

COUT3

COUT4

4.7 µF
100 V

4.7 µF
100 V

4.7 µF
100 V

100 nF
100 V

VIN

DR

LED5

A6271-1
SG
SP

PWMIN

GND

COMP

DITH

OSC

IREF

GND

22 nF
16 V

ROSC

RDITH
100 kΩ

CIREF

73.4 kΩ

2N7002
(DNP)

RSEN1

X2B

RSEN2

7.5 V Zener

LB

RC1
27 Ω

HB

CC1

CC2

470 nF
16 V

16 V

LED7
LED8

RLB2
10 kΩ

DZLB

SWLB

RIREF120 kΩ

RIREF2
10 kΩ
(DNP)

RPD
10 kΩ

RLB3
10 kΩ

Control Input for IREF
LB_cn

10 kΩ (DNP)
QIREF

LED6
Low Beam Enable Circuitry

MOS Current Sense Resistors(100 mΩ × 2)

VREG

RIREF3

SW

EPAD

10 nF
25 V
Additional for
LED Current Dimming

RSG 10 Ω
RSP 1.2 kΩ

RLB5

10 kΩ

RHB1
10 kΩ
RHB2
4.7 kΩ

1

RLB1
0.25 Ω

LED9
LED10

CSLW
47 nF, 50 V

3

CPWM

LED4

OVUV

SQJ46EP

VREG

LED3

LED Current Sense Resistors(1 Ω × 2)

LP
OVUV

LED2

RLED2

FAULTn

U1

RLED1

LED1

SPD15P10PLG

RFLT
2.7 kΩ

2

33 µH, 3 A

SWLB1

SWHB

2
2N7002

CREG

GND

1

RLB4
4k7

2N7002

VREG

VAN

DSW

LSW

SWOVP

ROV4
4.7 kΩ

100 nF 50 V

CBYP

15.4 kΩ

ROV3

LB

X1A

DOV1
VIN

DIN1

2N7002

HB

X1B

LB_cn
R3 Set for 350 kHz

Figure 22: Schematic of the automotive headlamp application.

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SCHEMATICS AND BILL OF MATERIAL
Figure 22 shows the developed schematic for the high-voltage
buck-boost converter driving 10 LEDs in high-beam and 7 LEDs
in low-beam with 400 mA LED current. The schematics also
includes a complete circuit of the high-beam to low-beamtransition with control from the 12 V battery input voltage.
Additional overvoltage protection circuitry is provided for theprotection of the driver in the scenario of an open-LED fault. A
list of the complete bill of material (BOM) for the developedboard is provided in Table 1.

CONCLUSION
The design and implementation of a high-voltage buck-boost
converter using A6271-1 is presented in this application note.
The following details are covered in this report:
1. Design details with a selection of various components.

APPLICATION INFORMATION

2. Steady-state performance of the driver on high-beam/ low-beam operation with details on thermal performance.
3. Transient behavior during the high-beam to low-beamand low-beam to high-beam operation with the impactof the slew rate control and selection of slew control
parameters.
4. Various protection schemes and implementation.
5. Developed schematics and the bill of material (BOM).

REFERENCES[1] A6271-1, Automotive High-Current LED Controller, https://www.allegromicro.com/en/products/regulate/led-drivers/leddrivers-for-lighting/a6271-1, Allegro MicroSystems, July 2019.
[2] Design Tool for High Voltage Buck-boost Converter Using
A6271-1, Allegro MicroSystems, December 2021.

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AN296293
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APPLICATION INFORMATION

Table 1: Bill of Material for Automotive Headlamp Application.
SectionDesignator
D IN1, D IN2
L IN

Power Input

1

Input Filter Capacitor, 4.7 µF, 50 V

3

C IN2, C IN5

Input Filter Capacitor, 100 nF, 50 V

2

TVS

50 V TVS for Load Dump Protection

1

FB1

Surge Current Fuse

1

U1

A6271-1 Power IC

1

C BYP

Bypass capacitor for Input Voltage, 100 nF, 50 V

1

C REG

Regulator output capacitor 1 µF, 16 V

1

R OSC

Oscillator resistor for PWM frequency set, 73.4 kΩ

1

R DITH

Dither Resistor, 100 kΩ

1

R C1

Comp pin for stabilization, 27 Ω

1

C C1

Comp pin for stabilization, 470 nF, 16 V

1

C C2

Comp pin for stabilization, 22 pF, 16 V

1

C IREF

IREF pin filter capacitor. 22 nF, 16 V

1

Provision for reference current control, DNP

0

R FLT

Fault pin pull-up resistor, 2.7 kΩ

1

C PWM

Filter capacitor on the PWMIN pin, 10 nF, 25 V

1

L SW

Switcher Inductor, 33 µH, 3 A

1

D SW

Buck-boost converter diode, MBRS2100

1SW

Buck-boost converter switch, SQJ46EP

1

R SEN1, R SEN2MOSFET current sense resistors, 100 mΩ

2

R SG

Switcher gate source resistor, 10 Ω

1

R PD

Switcher gate pulldown resistor, 10 kΩ

1

R SP

SP node current limit resistor, 1.2 kΩ

1

R ESD

ESD protection for LN pins, 150 Ω

2

C SEN

Filter capacitor for current sense 47 nF, 16 V

1

Output Capacitor, 4.7uF, 100 V

3

Output Filter Capacitor after Rled, 100 nF, 100 V

1

LED current sense resistor, 1 Ω each2P-channel bypass MOSFET, SPDP15P10PLG

1

Driving Transistors, 2N7002

2Power resistor, 0.25 Ω

1

R LB2, R LB3, R LB5, R HB1

gate drive resistors, 10 kΩ each4

R LB4, R HB2

C OUT4
R LED1, R LED2
SW LB
SW LB1, SW HB
R LB1

OVP

2

Filter Inductor, 30 µH, 4 A

C OUT1, C OUT2, C OUT3

LB/HB Circuit

Quantity

C IN1, C IN3, C IN4

R IREF1, R IREF2, R IREF3, Q IREF

A6271
Comp.

Description
Input Reverse Protection Diodes (100 V, 2 A)

gate drive resistors, 4.7 kΩ each

2

C SLW

Gate drain slew capacitor, 47 nF, 50 V

1

D ZLB

Gate pin protection Zener diode, 7.5 V

1

D OV1

Biasing diode, US1D

1

R OV1

Biasing resistor, 10 kΩ

1

D ZOV

Zener diode for OVP, 91V Zener

1

R OV2

Voltage divider for OVP, 4.2 kΩ

1

R OV3

Voltage divider for OVP, 15.4 kΩ

1

SWOVP

Driving transistor for OVP, 2N7002

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AN296293
MCO-0001439, Rev. 1

APPLICATION INFORMATION

Revision History
NumberDate

Description–

August 28, 2023

Initial release1

September 5, 2023

Updated introduction (page 1), Figure 3 (page 2), Equations 6, 7, and 9 (page 3), Equations 10 and 12 (page 4),
and minor editorial updates

Copyright 2023, Allegro MicroSystems.

The information contained in this document does not constitute any representation, warranty, assurance, guaranty, or inducement
by Allegro to the customer with respect to the subject matter of this document. The information being provided does notguarantee that a process based on this information will be reliable, or that Allegro has explored all of the possible failure modes.
It is the customer’s responsibility to do sufficient qualification testing of the final product to ensure that it is reliable and meetsall design requirements.
Copies of this document are considered uncontrolled documents.

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