DEVELOPMENT OF ELECTRICALLY SMALL PLANAR ANTENNAS WITH
MATCHING CIRCUIT
Haruichi Kanaya, Ryusuke Nabeshima, Ramesh K. Pokharel, and Keiji Yoshida
Graduate School of Information Science and Electrical Engineering, Kyushu University
744, Motooka, Fukuoka, 819-0395, JAPANkanaya@ed.kyushu-u.ac.jp
Abstract: We designed and fabricated an electrically small antenna (ESA) with coplanar waveguide (CPW)
matching circuit. Matching circuit is realized by using interdigital gap and transmission line. We designed
ESA with the aid of the commercial three-dimensional electro magnetic field simulator. We also madeexperiments on the ESA with CPW matching circuits using patterned circuit board.

1. INTRODUCTION
In micro- and millimeter- wave deices, integrating
entire transceivers on a single chip is the vision for
future wireless systems such as PDC, wireless LAN,
RF-ID and MIMO systems [1]-[3]. This has thebenefit of cost and size reduction.
However,antennas are considered to be the largest componentsof wireless systems, so that, it is necessary tominiaturize antennas [4].
Studies are also made of an electrically small
antenna (ESA), i.e., the antenna whose dimension ismuch smaller than one-wavelength, towards further
reduction of the antenna size [5]. It is widely
known, however, that in order to realize theminiaturized antenna, we must simultaneously realize
a broadband impedance matching circuit, which
compensates the narrow bandwidth peculiar to thesmall antenna with low radiation resistance, and wemust attain large impedance-matching ratios toconnect with semiconductor amplifiers with highinternal impedances. Also, the small antenna issensitive to the conductor resistance because of its
low radiation resistance, and the decrease of theradiation efficiency often makes serious problem.
The interest for the coplanar waveguide (CPW)transmission lines has increased significantly inrecent years. CPW transmission lines have lower
radiation leakage and less dispersion than microstrip
lines. Also, they are preferable for monolithic
microwave integrated circuit (MMIC’s) and RFIC
(radio frequency integrated circuit) since no via holes
are required for integration with devices [6, 7].
Slot antennas are suitable for CPW-fed
configuration, and the conventional CPW-fed slot
antenna is a one-wavelength center- fed type [8].
In our previous works, we designed the slot dipole
antenna whose length is one-wavelength with abandpass filter using CPW lines, which acts as an
impedance matching circuit as well [9-11]. In orderto reduce the total size of antenna, the slot loop
antenna was also designed whose length is half
wavelength in perimeter, and it was integrated to a
low noise amplifier with the matching circuit forinterconnecting them [12].
Also, we have designed a slot dipole antenna
whose dimension is much smaller than one
wavelength with aid of EM-simulator and carried out
the experiments on the slot dipole antenna with
2-poles bandpass filter, whose total size is 4.1 mm x
3.7 mm, using high temperature superconductors
YBCO thin film on the MgO substrate with relative
permittivity of 9.6, in the 5.0 GHz band [13].
In this paper, we design the impedance matchingcircuit, which connects an ESA fabricated on thenormal metal to a semiconductor amplifier. The
proposed matching circuits have performances
similar to those of the n-pole bandpass filter (BPF)[14]. By using the quarter wavelength transmission
line and admittance (J) -inverter, we can reduce thetotal antenna size more than one-wavelength. In the
beginning, theoretical performances of the ESA with
the CPW impedance matching circuit are calculated
by using the transmission line model as well as thecommercial electro-magnetic (EM) field simulator(Ansoft; HFSS ver.10). In order to demonstrate thetheory, we also carried out experiments on the ESA at
2.4 GHz, which has the size of 22.3 mm x 8.204 mm,with the pole number of n = 1 matching circuits using
FR4 printed circuit board.
2. DESIGN THEORY OF THE
ELECTRICALLY SMALL ANTENNA (ESA)
The conventional slot antenna is a one-wavelength
center feed slot antenna as shown in Fig. 1. The
designed center frequency is 2.4GHz-band. The
substrate has dielectric constant εr=4.25 and tanδ =0.015. The thickness of the substrate and copper
top metal is 0.8 mm and 18 μm, respectively.
Three-dimensional EM simulator simulates the RF
properties. Fig. 2 shows the return loss of thestandard slot antenna. The input impedance isalmost 50 Ω around 2.4 GHz, and –10 dB bandwidthis 165 MHz.
Fig. 3 shows the simulated radiation pattern of thestandard slot antenna shown in Fig. 1. The peak
gain and radiation efficiency are 3.192 dBi and96.07 %, respectively, which is the characteristics ofdipole antenna.

74.0mmW
feed⎧ L = 38 . 0 mm
⎪W = 18 . 0 mm⎪
Cross section⎨⎪ feed = 5 . 00 mm⎪⎩ edge = 1 . 00 mm 0.018mm

copper0.80mmY
Z

FR4: εr=4.25
tanδ=0.015(@1.8GHz)

X

22.30mm(0.18λ0)edge

8.204mm(0.067λ0)

24.0 mmL

metal loss, and Xa is the reactance of the antenna.
Because the quarter-wavelength parallel resonanceappears at 2.74 GHz, the size of this antenna is
electrically smaller than quarter wavelength when theantenna works at 2.4GHz. Za is 42.5+j385 Ω at 2.4
GHz, which is far from 50Ω, so that, return loss isalmost 0 dB. It is shown that most RF signals
reflect for the impedance miss-match.

Y

Za=Ra+jXa

Fig.1. Layout of the standard slot dipole antenna.

Z

X

0

Fig. 4. Layout of the ESA.

-10

4000

-15

Antenna Impedance (Za) [Ω]Rreturn Loss [dB]

-5

-20
-25
-30
-35
2

2.2

2.4
2.6
Frequency [GHz]

2.8

3

(a)

3000

Ra
Xa

2000
1000
0

-1000

Fig. 2. Return loss of the standard slot dipole
antenna.

-2000

2

2.2

2.4
2.6
Frequency [GHz]

2.8

3

2.2

2.4
2.6
Frequency [GHz]

2.8

3

0

θ

θ

-1

φ =0°

X

φ

Y

φ =90°

Fig. 3. Simulated radiation pattern of the standard
slot dipole antenna.

Return Loss [dB]

Z

(b)
-2
-3
-4
-5

2

Fig. 5. Input impedance (a) and return loss (b)of the ESA without matching circuit.
Fig. 4 shows the simulated layout of an electricallysmall antenna (ESA). The physical properties of thesubstrate are the same as those of standard antenna.
The antenna size is 0.18λ0 and 0.067λ0, respectively,
where λ0 is the wavelength in the vacuum at 2.45
GHz, which is especially smaller than that of thestandard dipole slot antenna. Figs. 5(a) and 5(b)show the Za (Za=Ra+jXa) and return loss of theantenna, where Ra represents radiation resistance and3. DESIGN THEORY OF IMPEDANCEMATCHING CIRCUIT BETWEEN ESA ANDSEMICONDUCTOR AMPLIFIER
In order to connect between ESA and RF front-end,
impedance matching must be attained between theantenna and amplifiers.

The present matching circuit is based on the
bandpass filter (BPF) composed of the transmission
line and J-inverters. Fig. 6(a) shows the n=1 BPF,
where, Y0 is the admittance of the load and B1 is thesusceptance in the parallel resonator with asusceptance slope parameter b1. The design
parameters of the n=1 BPF are
J 01 = w

Y0 b1
g 0 g1J 12 = w

b1Y0
g1 g 2(1)

n=1. In this figure, Za (=1/Ya) denotes the input
impedance of ESA and ZL (=1/YL) represents the
input impedance of LNA or the output impedance of
PA. Z01 (=1/Y01) and θi are characteristic impedanceand electrical length of the transition line. Fig. 7(b)shows the equivalent circuit model of the Fig. 7(a) at
the center frequency. In Fig. 7 (a), Y’ is theadmittance for looking the antenna from A-A’, andY’in is the admittance for looking amplifier from A-A’
and given by,(2)⎛ ω ω0 ⎞⎟⎟−B1 = b1 ⎜⎜⎝ ω0 ω ⎠(3)

where, w is the relative bandwidth and gi is the filter
parameter. Fig. 6(b) shows the equivalent circuit
model of the Fig. 6(a) at the center frequency. In
the figure, G’S and G’L, and Qei are the equivalentconductance and external quality factor, respectively.

Y' = G '+ jB ' = Y01

Ya + jY01 tan θ 01
Y01 + jYa tan θ 01

(4)

1
+ jY01 tan(−θ 02 )Z L + 1 / j ωC m
Yin ' = Gin '+ jBin ' =1
Y01 + jtan(−θ 02 )Z L + 1 / j ωC m

. (5)Finally, the comparison of the Fig. 6(b) and 7(b),the proposed design value θ01, Z01, θ02, Cm are led in
the numerical value as follows:(a)B' ω =ω = 0 (θ 01 ≤ π 2)

(6)b'
⎛ g g ⎞
= Qe1 ⎜ = 0 1 ⎟
G'
w ⎠
⎝

(7)

Bin ' ω =ω = 0 (θ 02 ≤ π 2)(8)b'⎛ gg ⎞= Qe 2 ⎜ = 1 2 ⎟
Gin '
w ⎠
⎝

(9)

0

Y0

jB1

J0,1

Y0

J1,2

(b)
G s′ =b1
Qe1

G L′ =

jB1

b2
Qe 2

0

Fig. 6. Circuit model of the n=1 bandpass filter(a) and equivalent circuit model at center
frequency (b).
Y’

A

Yin’

Antenna

Zin

Za

B

ZL

θ01
θ0

(b)
G’

A’

jB’

B’

−θ02

.

(10)

0

Amp.

Cm
Z01

ω 0 ∂B' ( Z 01 ,θ 01 )
.
∂ω
2
ω =ω

Fig. 8 shows the simulated layout of the ESA with
CPW transmission line which has Z01 and θ01 (See
A-A’ in Fig. 7 (a)). Fig. 9 shows the frequency
dependence of the Y’. We can see the parallel
resonance around 2.4 GHz.

22.30mmYin’

Fig. 7. Circuit model of the n=1 impedance matchingcircuit (a) and equivalent circuit model at center
frequency (b).

Fig. 7(a) shows the equivalent circuit model for thesmallest matching circuit, corresponding to BPF with

8.204mm

(a)

b' =

Y
Z

A

A’
Y’

X

Fig. 8. Layout of the electrically small antenna
with CPW transmission line.

0

3
2
1
0
-1

-10
-15
-20
-25
-30

-2
G'
B'

-3
-4

(b)

(a)

-5
Return Loss [dB]

Input admittance (Y') [mS]

4

2

2.2

2.4
2.6
Frequency (GHz)

2.8

-35

3

Fig. 9. Input admittance (Y’) of the ESA with
CPW transmission line.
Fig. 10 shows the layout of ESA with CPW matchingcircuit. In order to realize θ02 and Cm, we adopted
the interdigital gap and CPW transmission line,
where ZL is assumed to be Z0=50 Ω by theexperiment to be convenient. Fig. 11 shows the
input impedance (Zin), which is the impedance forlooking the antenna from B-B’ in Fig. 7 and Fig. 10.
Fig. 12 shows the return loss and Smith chart of the
ESA with CPW matching circuit. Zin is almost 50Ω
around 2.4 GHz, so that, return loss is –32.58 dB at
2.4 GHz. Fig. 13 shows the simulated radiation
pattern of the ESA with CPW matching circuit.
22.30mm (0.18λ0)

Layout8.204mm(0.067λ0)

L1
L2
W
B

1mm
InverterB’
Zin⎧ L1 = 10.0mm⎪ L = 4.60 mm⎪ 2⎪⎪W = 1.00 mm
J-Space⎨⎪ J − Space = 0.300 mm
Tooth Length ⎪⎪ToothLength = 0.200 mm⎩⎪ToothWidth = 0.148 mm
Tooth Width

Fig. 10. Layout of the ESA with CPW matchingcircuit.

2

2.2

2.4
2.6
Frequency [GHz]2.8

3

Fig. 12. Return loss (a) and Smith chart (b) of the
ESA with CPW matching circuit.

Z

θ

θ

φ =0°

Y

φ
φ =90°

X

Fig. 13. Simulated radiation pattern of the ESA.

4. EXPERIMENTAL
An ESA is fabricated on FR4 substrate by using
the print board making equipment (MITS; FP-21T
model 40), which has 100 μm-diameter milling cutter.
Fig. 14 shows the photographs of the antenna
fabrication system. In the figure, the close up of thehigh-frequency milling cutter is also shown.
Fig. 15 shows the photographs of the ESA with
CPW matching circuit. In the figure, an interdigital
gap is also shown. RF signal is input through
MMCX connecter, which has characteristic
impedance = 50 Ω. Fig. 16 shows the photographsof the RF measuring system. We measured the
S-parameters by using a GP-IB controlled vector
network analyzer (HP; HP8722C).

Input Impedance (Zin) [Ω]300

Printed board making equipment

Re[Zin]200

Im [Zin]
100
0

PCB

-100

High-frequency
Milling Cutter

-200
-300

2

2.2

2.4
2.6
Freqency [GHz]2.8

3

Fig. 11. Input impedance of the ESA with matchingcircuit.

Fig. 14. Photographs of the antenna fabrication
system.

result, we designed a small-size (22.30 mm x 8.204mm) slot antenna. Moreover, ESA with CPW
matching circuit was fabricated and measured the RF
properties, thereby we demonstrated frequency
characteristics as expected.
Inverter

0-5

Fig. 15. Photograph of the ESA.

Return Loss [dB]

Exp.
-10

Sim.

-15
-20
-25
-30

PC: Agilent
ADS
Vector Network
Analyzer:HP8722C

-35

2

2.2

2.4
2.6
Frequency [GHz]2.8

3

Fig. 17. Experimental result of the ESA.

VNA

0

MMCX-SMA
Connector

Fig. 16. Photographs of the RF measuring system.

5. RESULTS AND DISCUSSIONS
Fig. 17 shows the experimental results of the
return loss of the ESA with CPW matching circuit.
In the figure, broken line shows the return loss of the
EM simulated result. It seems that the bandwidthslightly decreases, and the center frequency shifts 37
MHz to lower side. Also, the return loss at thecenter frequency is –18 dB.
The observed
experimental results are caused by an error in edge
part of the interdigital gap (see Fig. 15), an error ofthe dielectric constant of the FR4 substrate andresidual loss of the connection between connector
and antenna.
Fig. 18 shows the comparison
between experimental results and simulated results
taking account of the permittivity error. In EM
simulation, we made the εr from 4.25 to 4.05, so that,the experiment results and the simulation values arecorresponding well.
Fig. 19 shows the comparison of the standard slot
antenna with the ESA. The antenna size can be
reduced to about 90 % and bandwidth design
becomes possible.
6. CONCLUSIONSIn this paper, slot dipole antenna with a bandpass
filter have been designed and tested. We succeededin realizing the circuit which matches the small
radiation resistance of ESA to the amplifier. As a-5
Return Loss [dB]

Antenna

Exp.

-10
-15
-20
-25
-30

Sim.

-35
2

2.2

2.4
2.6
Frequency [GHz]2.8

3

Fig. 18. Experimental results and simulated
results by taking account of permittivity error.

Fig. 19. Comparison of the standard slot antenna
with the ESA.
ACKNOWLEDGEMENT
This work was partly supported by a Grant-in-Aid
for Scientific Research (C) from the Japan Society

for the Promotion of Science (JSPS).
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