IEICE TRANS. COMMUN., VOL.E97–B, NO.7 JULY 2014

1395

PAPER

Decoupling Network Comprising Transmission Lines and Bridge
Resistance for Two-Element Array Antenna
Shumo LI†a) , Student Member and Naoki HONMA† , Member

SUMMARY
This paper presents a novel decoupling network consisting of transmission lines and a bridge resistance for a two-element array
antenna and evaluates its performance through simulations and measurements. To decouple the antennas, the phase of the mutual admittance between the antenna ports is rotated by using the transmission lines, and apure resistance working as a bridge resistance is inserted between the twoantenna ports to cancel the mutual coupling. The simulation results indicate that the proposed decoupling network can provide a wider bandwidththan the conventional approach. The proposed decoupling network is implemented and tested as a demonstration to confirm its performance. The
measurement results indicate that the mutual coupling between the two antenna ports is lowered by about 47 dB at the resonant frequency.
key words: decoupling network, mutual coupling, transmission lines,
bridge resistance, array antenna

1.

Introduction

Demand has been growing for the next-generation wireless communication system to oﬀer high-speed transmission
within the limited frequency resources. On the other hand,the space for the base-station antennas is limited, too, andsharing antenna space among several wireless systems is required. For such reason, several antennas need to be locatedimmediately adjacent to each other. In this case, the transmitted signal from one wireless system interferes with another one because of the high mutual coupling between theantennas even when their frequency bands are diﬀerent [1].
Since mutual coupling between the antennas becomes highwhen the multiple antennas are closely spaced, the communication performance is degraded. Also, when performing
power transmission and signal reception simultaneously inwireless power transmission systems, the strong transmitted signal leaks into the receiving antenna due to the mutual
coupling between the transmitting and receiving antennas;
this saturates the receiver and degrades communication performance. Therefore, high isolation over a broad band isrequired between the antenna elements.
The most common approach to reducing the eﬀects ofmutual coupling is to use a decoupling network (DN). Many
studies on DN have been published [2]–[11]. As indicated in[3], although a conventional decoupling and matching network (DMN) is eﬀective in suppressing the mutual coupling
between the antenna elements, the configuration of the cirManuscript received October 25, 2013.
Manuscript revised February 20, 2014.
†
The authors are with the Graduate School of Engineering,
Iwate University, Morioka-shi, 020-8551 Japan.
a) E-mail: t2312034@iwate-u.ac.jp
DOI: 10.1587/transcom.E97.B.1395

cuit is complex and its bandwidth is narrow. A DN designthat uses a microstrip line (MSL) as the susceptance has
been proposed; it oﬀers a simple circuit configuration [4].
However, if the mutual coupling between the antenna elements is small, the MSL must have such high impedance
that it cannot be realized since the line width would be narrower than the fabrication limit. The work [5] also presents
a design method of a simple decoupling and matching feeding network (DMFN) using transmission lines and bridge
susceptances. Only the design formulas of the DMFN havebeen derived, and how much the bandwidth can be obtainedby using this method has been not studied. As describedabove, a simple DN that is easy to realize and can suppress
the mutual coupling over a wide bandwidth is needed.
This paper proposes a novel DN comprising transmission lines and a bridge resistance for a two-element array
antenna. The DN uses the transmission lines to realize phaserotation and so address antenna mutual admittance while
the bridge resistance obtains the decoupling eﬀect [12], [13].
The result is excellent decoupling characteristics over a relatively wide bandwidth, even when antenna element spacing
is tight.
The next section describes the theory of the proposal.
The proposal is subjected to simulations and experiments toconfirm its wide bandwidth characteristics and its eﬀectiveness.
2.

Proposed Circuit Design

2.1 Proposed Decoupling Network Configuration
Figure 1 shows the configuration of the proposed DN. Here,Y a is the admittance matrix of the antenna. Y f is the admittance matrix of the transmission lines used to tune thephase of the mutual admittance of the antennas. The lengthand characteristic impedance of these lines are defined as l f
and Z f , respectively. As shown in Fig. 1(a), a pure resistance, R, is inserted as a bridge resistance. Since the two antenna elements are spatially separated, the two bridge MSLs
with characteristic impedance Z0 are used to connect thebridge resistance. When the bridge MSLs have line lengthsof λg /2 (λg : eﬀective wavelength), the equivalent model ofthe bridge circuit is simplified to a purely resistive element.
The admittance matrix of the bridge circuit comprising the
MSLs and resistance is defined as Y d . Figure 1(b) shows
an equivalent circuit of the DN. The details of the derivation procedure of Figs. 1(b) from (a) will be described in thec 2014 The Institute of Electronics, Information and Communication Engineers
Copyright IEICE TRANS. COMMUN., VOL.E97–B, NO.7 JULY 20141396

MSL is obtained asY  = Y f 11 − Y f 12 (Y a + Y f 22 )−1 Y f 21 .

(2)

For lossless transmission lines, Y f , is determined by the line
length, l f , and the characteristic admittance, Y f . Additionally, Y f 11 , Y f 12 , Y f 21 , Y f 22 are given by
jY f cot βl f0
Y f 11 = Y f 22 = −(3)0
jY f cot βl f
jY f csc βl f0
Y f 12 = Y f 21 =,(4)0
jY f csc βl fwhere β is wave number in the phase rotation MSL. By expanding (2), the MSL length, l f , which makes the imaginary
part of y21 (y21 is the oﬀ-diagonal elements of (2)) zero, isobtained by
2A1
l f = tan−1
.
(5)
√
β
−B ± B2 − 4AC
Here, A, B, and C are described as
A = Y 2f ya12i
B = −Y f [ya12r (ya11r + ya22r ) + ya12i (ya11i + ya22i )]C = ya12r (ya11r ya22i + ya22r ya11i − ya12r ya21i )− ya12i (ya11r ya22r − ya11i ya22i + ya12i ya21i ),Fig. 1

Proposed decoupling network model.

(6)

where ya11r , ya12r , ya21r , ya22r are the real part of each element of Y a , and ya11i , ya12i , ya21i , ya22i are the imaginary
part of each element of Y a , respectively. At this time, theresistance value, R, is obtained from R = 1/y21 . The bridge
resistance is connected between the antennas to achieve decoupling, so the admittance matrix of the DN is represented
by

 

yb −yb
1/R −1/R
Yd =
=
.
(7)
−yb yb
−1/R 1/R

next Sect. 2.2. In Fig. 1(b), y12 is the mutual admittance; it
represents the mutual coupling between the antennas. Whenyb = y12 , the mutual admittance between feeding port 1 andport 2 would be zero which cancels mutual coupling completely. Since R = 1/yb , y12 must be real positive value to be
successfully cancelled by the bridge resistance. Therefore,
the length, l f , of the MSL for phase rotation is determinedso as to transform y12 into a real positive value.

Since the DN is connected in parallel to the antennas, theobserved admittance matrix Y  with decoupling is denotedas

2.2 Theory for Circuit Design

From (8), it can be seen that decoupling is realized because
the oﬀ-diagonal elements of Y  are zero.

A method of deriving circuit design values is described.
First, the admittance matrix of the antennas, Y a , is evaluated. Then the MSL length, l f , and the resistance value, R,
are calculated by using the following formula. The admittance matrix of the MSL for phase rotation, Y f , is split into
four partitioned matrices asY f 11 Y f 12
Yf =
,
(1)
Y f 21 Y f 22
where the subscript numbers represent the reference planes
shown in Fig. 1(a). When Y f is connected to Y a , the observed admittance matrix at the end of the phase rotation

Y  = Y  + Y d .

3.

(8)

Simulation

3.1 Simulation Conditions
To confirm that the proposed DN oﬀers wide bandwidth,
we simulated the decoupling circuit characteristics and compared to DN with the capacitance instead of the resistance.
Here, the DN with capacitance is achieved by replacing theresistance to capacitance in Fig. 1, and the MSL length, l f , isdetermined so as that the real part of y21 becomes zero. Thissection starts be describing the simulation conditions. Figure 2 shows the simulated model of the two basic antennas.

LI and HONMA: DECOUPLING NETWORK COMPRISING TRANSMISSION LINES AND BRIDGE RESISTANCE FOR TWO-ELEMENT ARRAY ANTENNA

1397

Fig. 2

Simulated model of antennas.

In this study, both antennas are monopoles with resonant
frequency of 2.4 GHz. The element spacing, d, between theantennas is varied by 0.1 λ0 (λ0 : wavelength in vacuum)from 0.1 λ0 to 1.0 λ0 . The circuit characteristics are evaluated for each element spacing.
3.2 Simulation Results
Figure 3 plots simulated S -parameters versus the element
spacing between the antennas at the resonant frequency.
Here, the resonant frequency is defined as the frequency
where the reflection, S 11 , of the element alone becomes minimum. Since S -parameter is symmetric in this model, onlyreflection, S 11 , and mutual coupling, S 21 , are shown. In
addition, the coupling characteristics without DN and thesimulated DN with capacitance are also shown here. Figure 3(a) confirms that the reflection characteristic, S 11 , of
decoupling with capacitance is the same as that with resistance. The reflection characteristics of the antenna itself are
between −10 dB and −25 dB, and the reflection characteristics of the antenna with DN are between −10 dB to −20 dB,
too. Although the reflection level, S 11 , with DN is higher
than that without DN, there is no significant eﬀect of DN on
the antenna performance since S 11 is still lower than −10 dB
with various d. The degradation in S 11 with decoupling isthought to be due to the addition of the diagonal elements of
Y d that cause changes in self-admittance and the eﬀect on
the reflection coeﬃcient. In Fig. 3(b), the mutual coupling,
S 21 , is simply reduced as antenna element spacing, d, is increased. Here, only the raw S 21 value without DN is shownsince S 21 with DN is always 0. This result shows that the
proposed DN yields high isolation between the antennas regardless of antenna element spacing. Also, from this result,it can be seen that mutual coupling still remains even with
the increase of the distance between the antennas. This canbe said that the DN is certainly needed to obtain high isolation of the antennas even if the antenna element spacing isincreased.
Figures 4(a) and (b) show the frequency characteristics
of mutual coupling, S 21 , for the antenna element spacing of
0.2 λ0 and 0.8 λ0 , respectively. The cases with resistanceFig. 3S -parameters versus antenna element spacing.

and capacitance were simulated. From Figs. 4(a) and (b),it can be seen that the mutual coupling, S 21 , were successfully lowered by DNs with both capacitance and resistanceregardless of the antenna element spacing, d. Also, it can beseen that the DN with resistance suppresses the mutual coupling, S 21 , over the wide band of 2.1 GHz∼2.7 GHz whend = 0.2 λ0 . On the other hand, it is found that the decoupling
bandwidth for the DNs with the resistance and capacitanceis almost identical when d = 0.8 λ0 .
Figure 5 shows the fractional bandwidth versus antenna element spacing between the antennas. The bandwidth is defined as a frequency range where the mutual coupling is lower than a certain value. The evaluation of the−20 dB bandwidth versus antenna element spacing is shownin Fig. 5(a) and the evaluation of the −30 dB bandwidth versus antenna element spacing is shown in Fig. 5(b). For comparison, the results gained when using the capacitance arealso shown. Figure 5 shows that when the antenna element
spacing is narrower than 0.7 λ0 , using the resistance yields
relatively wider available bandwidth than using the capacitance. This result indicates that the proposed decoupling design is eﬀective, especially when the antenna element spac-

IEICE TRANS. COMMUN., VOL.E97–B, NO.7 JULY 2014

1398

Fig. 4Frequency characteristics of mutual coupling S 21 .

ing becomes narrow.
Figures 6(a) and (b) show the frequency characteristics of real and imaginary parts of the mutual admittance,y21 , respectively. Here, the mutual admittance is observedat the reference plane 1 shown in Fig. 1(a). Note that theline lengths, l f , in DNs with resistance and capacitance are
diﬀerent since the required degree of the phase rotation is
diﬀerent for these two DNs. The antenna element spacing
was set to d = 0.2 λ0 , 0.5 λ0 , 0.7 λ0 , and 1.0 λ0 as an example. Figure 6(b) shows that the curves for DN with capacitance are steeper than that with resistance when the antenna
spacing is narrow. Since y21 and the bridge admittance, yb ,should be cancelled out each other, the rapid change yields
narrow bandwidth of DN. Therefore, the bandwidth of the
DN with capacitance becomes narrow in this case. On thecontrary, the curves for DN with resistance in Figs. 6(a) and(b) are almost constant when d = 0.2 λ0 , and this means
the wide decoupling bandwidth is expected since y21 and yb
cancel out each other over the wide range. However, thespectral fluctuation of the mutual admittance, y21 , of DN
with the resistance becomes large when d > 0.7 λ0 . Thisaﬀects the bandwidth of the decoupling characteristics. For
the above reason, the DN with resistance loses the advantage

Fig. 5Fractional bandwidth versus antenna element spacing.

in bandwidth when d ≥ 0.7 λ0 .
Adding a resistance to the circuit generally increases
the insertion loss, which cannot be ignored. Figure 7 plots
the radiation eﬃciency of the DN with resistance and withcapacitance versus the antenna element spacing. The result
without DN is also shown. From Fig. 7, it can be confirmedthat the radiation eﬃciency of the DN with capacitance isclose to 100 percent regardless of the antenna element spacing. However, while the radiation eﬃciency of the DN withresistance increases with antenna element spacing, it is always lower than that with capacitance. This result showsthat the radiation eﬃciency of the proposed DN is reduced
if the resistance is used. Additionally, it can also be seen thatcompared to the result without DN, the radiation eﬃciency
of DN with capacitance falls as the antenna element spacing
exceeds 0.5 λ0 . This is because the radiation eﬃciency also
includes the reflection loss of the circuit.
Figure 8 shows the S -parameter model of the antenna
and DN. In Fig. 8, a1 ,a2 are the waves incident to the DN,
and b1 ,b2 are the waves reflected from the DN. The insertion
loss of the DN is calculated by


Pout
.
(9)
Loss = 10 log10
PinLI and HONMA: DECOUPLING NETWORK COMPRISING TRANSMISSION LINES AND BRIDGE RESISTANCE FOR TWO-ELEMENT ARRAY ANTENNA

1399

Fig. 8Fig. 9Fig. 6

Frenquency characteristics of mutual admittance, y21 .

Fig. 7

Radiation eﬃciency versus antenna element spacing.

Here, Pin is the input power to the DN, Pout is the input
power to the antennas; they are defined by
Pin = |a1 |2 − |b1 |2
Pout = |b2 |2 − |a2 |2 .

(10)

Figure 9 shows the insertion losses of the proposed

S -parameter model of antenna and DN.

Insertion losses versus antenna element spacing.

DN versus the antenna element spacing. Here, the bridge
resistance values were determined to suit the resonant frequency, 2.4 GHz. When the frequency range is defined as
2.1 GHz∼2.7 GHz temporarily, the results of the insertion
loss at a frequency of 10 percent and 30 percent value ofthe bandwidth are also presented. From Fig. 9, it can befound that the circuit loss is large when the antenna element
spacing is narrow if either at the resonant frequency or not.
The maximum circuit loss at any frequency indicated, under 3 dB, occurs at the narrowest antenna element spacing,d = 0.1 λ0 .
Figure 10 shows the impact of DN with resistance andwith capacitance on the xy-plane radiation pattern. Here, thexy-plane radiation pattern is shown as the actual gain; theresults without DN are also shown. Since the antenna structure is symmetric, only the radiation patterns when feeding
port 1 are shown. (a), (b), and (c) are for the antenna element
spacing values of d = 0.2 λ0 , d = 0.5 λ0 , d = 0.8 λ0 , respectively. By comparing the results with and without DN, it
can be seen that there is no significant change in the radiation patterns and gain with either resistance or capacitance.
In addition to that, compared the results with DN using theresistance and capacitance, the eﬀect on the antenna gain ismuch larger when the proposed DN is used. This is causedby the circuit loss of the proposed DN using the resistance.

IEICE TRANS. COMMUN., VOL.E97–B, NO.7 JULY 2014

1400

Fig. 11

Fig. 12Fig. 104.

Measurement setup.

Configuration of monopole antennas.

Impact of decoupling network on radiation pattern.

Experimental Demonstration

4.1 Design of Fabricated Circuit
In this section, the decoupling eﬀect of the proposed circuitdesign method is demonstrated by measurements on a twoelement monopole array antenna following the simulation
setup. Figure 11 shows the photo of the measurement setup.
The antenna element spacing, d, is 0.5 λ0 , i.e., d = 62.5 mm.
Figure 12 shows the monopole antenna configuration
designed for this experimental demonstration. The rectangle aluminum plate, whose dimensions are W x × Wy = 300 ×
300 mm2 , is used as a finite ground plane of the two-elementmonopole antenna. The monopole antennas are configured
symmetrically on the ground plate and are cylindrical conductor whose dimensions are given as L = 28.9 mm anda = 3 mm. The resonant frequency of the antennas is in thevicinity of 2.4 GHz.
Figure 13(a) shows the design of the proposed DN

Fig. 13

Design for proposed decoupling network.

based on measured values of the antenna elements. The values of line length, l f , of the MSL for phase rotation and thebridge resistance, R, were calculated by using the derivation
method mentioned in Sect. 2. Here, l f = 73.1 mm and R =113.4 Ω. Figure 13(b) is a photo of the fabricated DN. Theline length of the two bridge MSLs is l0 = 45.8 mm. The
two MSLs are bent to fit within the antenna element spacing
since total MSL length is greater than the element spacing of
the antenna. The bridge resistance was realized as a chip resistor with R = 110.1 Ω; this value is only slightly diﬀerent
from the desired value.

LI and HONMA: DECOUPLING NETWORK COMPRISING TRANSMISSION LINES AND BRIDGE RESISTANCE FOR TWO-ELEMENT ARRAY ANTENNA

1401

Fig. 15

Fig. 14Frequency characteristics of S -parameters.

4.2 Measurement Results
Figure 14 shows the frequency characteristics of the measured S -parameters. Here, the simulated DN and the coupling characteristics without the DN on simulation and experiment were also shown. It should be noted that the simulation results shown here were calculated by the measured
values of the antennas and the design values of the DN. The
self-admittance after decoupling is changed by using the resistance, and this can alter the reflection coeﬃcient. Becauseof this, both the simulated and the measured S 11 after decoupling are changed in Fig. 14(a). Comparing the simulated
and the measured S 11 , it can be seen the resonant frequency
is shifted. The reason of this can be considered to be dueto the inclusion of the imaginary component in the chip resistor used in the measurement system. From Fig. 14(b), it
can be seen that simulated and measured S 21 values after
decoupling generally agree. Also, it can be seen that the operation frequency is the same as the resonant frequency andthe mutual coupling, S 21 , can be suppressed by about 47 dB
at maximum.
Figure 15 shows the frequency characteristics of the insertion loss when using the proposed DN; the simulated re-Frequency characteristics of insertion loss.

sult is also shown. From this figure, it can be found that thecircuit loss due to the resistance is below 2 dB in both thesimulation and the experiment. Furthermore, the results ofcircuit loss shown in Fig. 9 and Fig. 15 are compared with at
the resonant frequency, 2.4 GHz. The simulated circuit loss
shown in Fig. 9 at 2.4 GHz is about 1.5 dB, the circuit loss
by calculation and measurement shown in Fig. 15 at 2.4 GHz
are about 1.51 dB and 1.86 dB, respectively. These results
show that there is no significant diﬀerence between the simulated and measured losses. According to the simulation
described in Sect. 3, the radiation eﬃciency is mainly determined by the DN since the radiation eﬃciency of the antenna elements except for the reflection and the circuit loss
is approximately 100 percent. Similarly, it is estimated thatthe radiation eﬃciency of the antennas with proposed DN
would be around 65 percent on the basis of the circuit loss
in experiment.
Figures 16(a) and (b) show the measured and simulated
radiation patterns for port 1 and 2, respectively, for DN andwithout DN. The radiation patterns are shown in the actual
gain where the reflection and the circuit losses are considered. Figure 16 confirms that the measured and simulated
radiation patterns are in quite good agreement. Since themeasured patterns and gain show no significant change after
decoupling, it can be said that the circuit loss of the proposed
DN has small impact on the radiation pattern and eﬃciency
of antenna.
5.

Conclusion

This paper proposed a decoupling network comprisingtransmission lines and bridge resistance for a two-element
array antenna. MSLs for phase rotation are connected in series to the antennas to transform the phase of the mutual
admittance, and the decoupling eﬀect is realized by connecting a bridge resistance in parallel between the antennas.
Simulations of the monopole array antennas showed that the
proposed DN using the resistance can obtain a wider bandwidth as compared to the DN using a capacitance when theantenna element spacing is narrower than 0.7 λ0 . Moreover,experiments showed that the mutual coupling is improved

IEICE TRANS. COMMUN., VOL.E97–B, NO.7 JULY 20141402

ements,” IEICE Communications Express, vol.1, no.7, pp.246–251,
Dec. 2012.
[5] K. Kagoshima, S. Obote, A. Kagaya, K. Nishimura, and N. Endo,“An array antenna for MIMO systems with decoupling network using bridge susceptances,” 2011 Wireless VITAE, 2011.
[6] S.C. Chen, Y.S. Wang, and S.J. Chung, “A decoupling technique forincreasing the port isolation between two strongly coupled antennas,” IEEE Trans. Antennas Propag., vol.56, no.12, pp.3650–3658,
Dec. 2008.
[7] A.C.K. Mak, C.R. Rowell, and R.D. Murch, “Isolation enhancement between two closely packed antennas,” IEEE Trans. Antennas
Propag., vol.56, no.11, pp.3411–3419, Nov. 2008.
[8] J.C. Coetzee and Y. Yu, “Port decoupling for small arrays by meansof an eigenmode feed network,” IEEE Trans. Antennas Propag.,vol.56, no.6, pp.1587–1593, June 2008.
[9] J. Itoh, N.T. Hung, and H, Morishita, “The mutual coupling reduction between two j-shaped folded monopole antennas for handset,”
IEICE Trans. Commun., vol.E94-B, no.5, pp.1161–1167, May 2011.
[10] H. Jiang, R. Yamaguchi, and K. Cho, “Reduction in mutual coupling
characteristics of slot-coupled plannar antenna due to rectangularelements,” 2008 IEEE Intl. ANt and Prop. Society Symp., 137.7 July
2008.
[11] H. Jiang, R. Yamaguchi, and K. Cho, “Bandwidth characteristic improvement of filter integrated antenna composed of aperture coupledpatch antenna by using trapezoidal element,” IEICE Trans. Commun., vol.E92-B, no.12, pp.3960–3963, Dec. 2009.
[12] S. Li, N. Honma, and N. Yamaki, “Proposal of tunable decoupling
network comprising transmission lines and lumped elements,” 2012
International Symposium on Antenna and Propagation (ISAP 2012),
Electroc Proc. ISAP 2012, 1E4-2, Oct. 2012.
[13] S. Li, N. Honma, and N. Yamaki, “Decoupling network comprising transmission lines and bridge resistance for two-element array
antenna,” 2012 Asia Pacific Microwave Conference (APMC2012),
3C4-04, Dec. 2012.

Fig. 16

Radiation patterns (d = 0.5λ0 ).

by about 47 dB at the resonant frequency without serious
degradation in the gain of the radiation patterns. These results prove that the proposed decoupling network is eﬀective
in suppressing the mutual coupling over a broad bandwidtheven if the element spacing is narrow.
Acknowledgements
This research was partially supported by JSPS KAKENHI(25709030).
References[1] A. Kishida, T. Hiraguri, M. Ogawa, K. Nishimori, N. Honma, and
T. Sakata, “A novel interference avoidance technique on mobile
wireless routers using IEEE802.11nPSMP,” IEICE Trans. Commun.,vol.E93-B, no.8, pp.2053–2062, Aug. 2010.
[2] J.B. Andersen and H.H. Rasmussen, “Decoupling and descattering
networks for antennas,” IEEE Trans. Antennas Propag., vol.AP-24,
no.6, pp.841–846, Nov. 1976.
[3] J. Weber, C. Volmer, K. Blau, R. Stephan, and M.A. Hein, “Miniaturized antenna arrays using decoupling networks with realistic elements,” IEEE Trans. Wicrow. Theory Tech., vol.54, no.6, pp.2733–
2740, June 2006.
[4] N. Yamaki and N. Honma, “Decoupling technique for n-element linear array antenna using transmission lines between neighboring el-

Shumo Liwas born in Changchun, Jilin,
China, in 1988 and came to Japan for study
in 2006. She received the B.E. and M.E. degrees in electrical and electronic engineeringfrom Iwate University, Morioka, Japan in 2012
and 2014, respectively. She joined NTT docomo
in 2014. Her research interest is decoupling network which applies to wireless power transmission using array antennas.

Naoki Honma
received the B.E., M.E., andPh.D. degrees in electrical engineering from Tohoku University, Sendai, Japan in 1996, 1998,
and 2005, respectively. In 1998, he joined the
NTT Radio Communication Systems Laboratories, Nippon Telegraph and Telephone Corporation (NTT), in Japan. He is now associate
professor in Iwate University. He is an Associate Editor for the Transactions on Communications for the IEICE Communications Society
from May 2011. He received the Young Engineers Award from the IEICE of Japan in 2003, the APMC Best Paper Awardin 2003, and Distinguished Service Award from the IEICE Communications Society in 2011. His current research interest is MIMO antenna technology for high-speed wireless communication systems. He is a memberof the Institute of Electronics, Information and Communication Engineers(IEICE) of Japan.