Selected Papers

Planar Beam-Scanning Microstrip Antenna Using Tunable
Reactance Devices for Satellite Communication Mobile Terminal
Naoki Honma†, Tomohiro Seki, and Koichi Tsunekawa
Abstract
A series-fed beam-scanning array using a multi-stage configured microstrip antenna with tunable reactance devices (MUSCAT) for satellite communication mobile terminals is proposed. This antenna significantly expands the beam scanning range and achieves high efficiency. It comprises unit element
groups, whose elements are placed close to each other and use tunable reactance devices. Analyses andexperiments on the unit element groups showed that their multi-stage configuration extends the phaseshift range and increases the radiation efficiency (e.g., a phase shift of 120° and radiation efficiency ofmore than 50%) when three stages are used. The radiation pattern of a fabricated MUSCAT array antenna with eight unit element groups was measured. The beam scanning range was 27°, which is more than
twice that of a non-multi-stage configuration.

1. Introduction
A mobile satellite communication system can beused to provide geographically seamless communication service even in a rural area without an infrastructure [1], [2]. Since this system requires terminal
antennas with high actual gain to obtain a high data
rate for downlink transmission, the mobile terminalmust constantly direct its beam toward the satellite.
To provide such satellite communication services fora mobile vehicle, which may move at high speed, wemust use a low-profile satellite-tracking antenna for
aerodynamic reasons.
The combination of a one-dimensional planar
phased array antenna and a rotator for azimuth control is one feasible solution. However, the configuration and control of the phased array antenna is complicated even for one-dimensional beam scanningbecause a lot of phase shifters are employed. Moreover, each phase shifter needs a low-noise amplifier
to compensate for the insertion loss of the phaseshifter. This causes high power consumption.
A series-fed beam-scanning array antenna can simplify the control of the phase shifters because series† NTT Network Innovation Laboratories
Yokosuka-shi, 239-0847 Japan
E-mail: honma.naoki@lab.ntt.co.jp

24

fed circuits enable the antennas to scan the beam
direction easily with only a single control signal forthe phase shifters. In addition, the insertion loss due
to conduction loss can be reduced because this antenna can have a short feedline. James et al. [3] reported
an example of a planar beam-scanning antenna
employing such a feed circuit and proposed a design
for an antenna employing varactor diodes as tunable
reactance devices inserted between the strip conductor and the ground plane. Since it uses a planar antenna on the substrate, all of the devices can be easily
mounted on the substrate. This planar array antenna
can be constructed compactly. The varactor diodes,
themselves, consume little power because little current flows into a varactor with reverse DC-bias. Whenvaractor diodes with low insertion resistance areused, there is no need to use many low-noise amplifiers. Although this antenna reduces the complexityand loss, the reactance of the devices must bechanged significantly when wide-range scanning isrequired. In this case, the reflection power and loss atthe devices accumulate in the series feedline, andthese factors affect the radiation pattern. This problem reduces the beam-scanning angle.
In this paper, we propose a novel series-fed beamscanning array antenna consisting of a multi-stage
configured microstrip antenna with tunable reactancedevices (MUSCAT). Each element group comprises
NTT Technical Review

Selected Papers

Feed port

Unit element group

Tunable reactance devices

SubstrateDz
RF-signal
DC-bias

θ

y
φ

d

d

x

Termination load
Microstrip line

Resonant direction

Microstrip antennas

Fig. 1. Geometry of MUSCAT array antenna.

several microstrip antennas placed close together anduses tunable reactance devices. This provides a lowprofile phased array antenna with a simple geometry
and low-loss beam-scanning performance. Numerical and experimental results for the phase shift performance and radiation properties of the unit element
group are presented. The beam scanning characteristics of a fabricated antenna with eight element groups
are also shown to demonstrate wide-beam scanningand easy operation with low loss.
2. Geometry of MUSCAT array antenna
The geometry of the MUSCAT array antenna isshown in Fig. 1. Each element group is constructed ofnarrow microstrip antennas connected to each otherby microstrip lines, where the element spacing d ismuch less than the wavelength in vacuum. The resonant direction of the narrow microstrip antenna isparallel to the long side of the rectangular antenna.
The microstrip lines at both ends of the microstrip
antenna are configured so as to cancel the undesiredradiation generated by themselves. Two reactancedevices, which can be tuned using DC-voltage, areattached to the ends of the microstrip antenna. Figure
2 is a cross section of the MUSCAT array. The
devices are connected to the microstrip antennas andthe ground plane via through holes.
The MUSCAT array comprises several unit element
groups and the spacing between neighboring groups
denoted by D is much greater than element spacing d.
All of the unit element groups have the same dimensions and are connected in series by microstrip lines
having an ‘S’ shape to adjust the phases in the unit
element groups. In this traveling wave array antenna,
some of the power incident to one antenna element isradiated and the remaining power is delivered to thenext element. Since the delay of the signal also propagates from one element to the next and accumulatesVol. 3 No. 9 Sep. 2005

Microstrip antennas
Substrate

Through hole

Tunable reactance device
Ground plane

Fig. 2. Cross section of antenna.

through the series feedline, it is easy to provide a relatively large beam scanning range without employing
360° phase shifters. Furthermore, this antenna canprovide large phase shifts in the unit element groups
because these groups have a multi-stage configuration, and the unit element groups, which all haveidentical dimensions, achieve a phase front without
distortion when all of the reactance devices are tuned
to the same state. The termination load connected tothe end of the array suppresses the reflection of thetraveling wave, which causes undesired radiation [4],[5]. This geometry can control all the reactancedevices with a single DC bias. The RF-signal and theDC bias are supplied through the same feedline. Thismechanism simplifies the feed and bias network configuration and enables easy beam steering.
3. Properties of unit element group
The MUSCAT array has a series-fed configuration.
However, the accumulated insertion loss and thereflection at the input port of the unit element groupin the series array antenna cause a reduction in theactual gain and produce a backward traveling wavethat generates undesired radiation. Moreover, manytunable reactance elements are used in the MUSCAT
array and the power consumed by them is not negligible. For these reasons, it is necessary to investigate
the radiation, reflection, and loss of the unit element
25Selected Papers

λ 0, and the distance between two varactor diodes in
the microstrip antennas is s. The thickness of the substrate is 0.02 λ 0, and the relative dielectric constant is
εr = 2.2. The length of the microstrip antenna is 0.39
λ g where λ g is the effective wavelength in the
microstrip antenna. The antenna length is obtained
from the effective dielectric constant given in [7]. The
operating frequency is 2.4 GHz.

150

2.5

100

2

50

1.5

0

1

–50

0.5

–100
0

5

10

15
20
DC-voltage (V)25

3.1 Numerical analysis of radiation and phaseshift performance in unit element group
We investigated the relationship between the number of elements m and the phase shift. The magnitude
and phase of S21 obtained from the calculation for m= 3 are shown in Fig. 5. Here, the distance between
two varactor diodes is set to 0.153 λ g. The ∆|S21| andΦ are defined as the transmission power difference
and phase difference from the reference S21 at 30 V,
respectively. The Φmax is the Φ at the DC-voltage
where the magnitude of S21 is 1 dB below the magnitude of S21 at 30 V.
The results shown in Fig. 5 indicate that the transmission power decreases as the DC-voltage decreases. This is due to the loss caused by the varactor
diodes when the DC-voltage is low, as shown in Fig.
3.
Next, we investigated the relationship between
spacing s of the two varactor diodes in the microstrip
antenna and Φmax of the unit element group. The calculated phase shift performance of the unit element
group is shown in Fig. 6. The characteristic impedance of the input or output feedline was optimized toobtain the largest Φmax, and the optimized value ofthe impedance Z 0 is shown in Fig. 6(b). There is a discontinuity in Z 0 around 0.16 λ g < s < 0.17 λ g, which

Resistance, R (Ω)

Reactance, X (Ω)

group in order to design the MUSCAT array.
To enable us to discuss the actual performance ofthe MUSCAT array, we measured the impedance ofthe varactor diodes. Figure 3 shows the measured
impedance of the varactor diode versus DC-bias voltage when a Renesas Technology HVU316 was usedas the tunable reactance device and the operating frequency is 2.4 GHz. Since high absolute values for thereactance and low resistance are observed in the highvoltage range, the bias voltage should be as high as
possible so as to decrease the loss in the varactor
diodes.
Based on these results, we investigated the scattering parameters of the unit element group numericallyusing the moment method [6] and experimentally. A
model of the unit element group is shown in Fig. 4,
where λ 0 is the wavelength in vacuum, m denotes thenumber of stages (i.e., the number of elements in theunit element group), the element spacing is d = 0.12

030

Fig. 3. Measured impedance of varactor diode versus
DC-voltage.
m

Varactor diodes
Port #2s
Port #1

0.39 λg(0.29 λ0)•••d
Microstrip antennas

Fig. 4. Configuration of unit element group.

26

NTT Technical Review

Selected Papers∆|S21| = –1 dB
0

130∆|S21|

120–1–Φmax (deg.)|S21| (dB)

110
–2
–3

100
90
80

–4
–5

(a)
5

10

15
20
Voltage (V)

25

60
0.12

30

0.13

0.14

0.15

0.16
s /λg

0.17

0.18

0.19

0.20

90

0
–50

80

Φmax

–100
–150

Z 0 (Ω)Φ (deg.)

(a)

70

–200

70

60

–250
50

–300

(b)

(b)

–350
10

15
20
Voltage (V)(m = 3)25

Fig. 5. Calculated |S21| and Φ versus voltage: (a) |S21|
and (b) Φ.

corresponds to the distance between the two feedlines
at the microstrip antenna. Considering this, we conjectured that this discontinuity is caused by the feedlines. The results in Fig. 6 show that the largest Φmax
value is obtained for s = 0.153 λ g. In the following
discussion, this optimum distance is used.
Figure 7 shows the relationship between the number of elements m and the maximum phase shift Φmax
obtained from the calculation. The Φmax increases
almost linearly as m increases. This means that theeffect of the mutual coupling between the elements inthe multi-stage configuration of the unit element
group has little effect on the phase shift performance.
To clarify the radiation performance in the unit element group, we must consider the transmission
power because it is incident to the next unit element
group and is radiated. In this paper, the radiation efficiency ηe of the unit element group is defined asVol. 3 No. 9 Sep. 2005

40
0.12

30

0.13

0.14

0.15

0.16 0.17s /λg(m = 3)

0.18

0.19

0.20

Fig. 6. Unit element group properties versus varactor
diode distance s: (a) Φmax and (b) characteristic
impedance of feedline Z 0.

250
200
–Φmax (deg.)

5

150
100
50
0

1

2

3

4

5

6

Number of elements, m

Fig. 7. Maximum phase shift Φmax versus number of
elements m.

27

Selected Papers

Prad
2 ,
Pinc (1 − S21 )(1)

where Pinc is the incident power at input port #1 of
one unit element group and Prad is the radiation
power of the unit element group. The calculated radiation efficiency of the unit element group obtained
from Eq. (1) is shown in Fig. 8. Here, the average ofthe radiation efficiency is used for simplicity because
ηe depends on the bias voltage. The ηe represents theaveraged radiation efficiency and is defined byηe = ∫

Φ max

0ηe dΦ / Φ max .

(2)

The results shown in Fig. 8 indicate that ηe increases with m. This is because the low loss range of thevaractor diode is used in the unit element group witha large m. Thus, the proposed configuration canachieve both a large phase shift and a high level ofradiation efficiency when the number of the elements
is large.
In the following discussion, m is set to 3 so that the
spacing between neighboring unit element groups is
D = 0.5 λ 0, since the unit element groups with m > 3
require a D value larger than 0.5 λ 0 and a grating lobe
is generated when the beam of the MUSCAT array issteered. In addition, more than 50% of the averageradiation efficiency ηe is obtained in the unit element
group with m = 3 and the increase in the radiation efficiency is not so large when m is greater than 3.
The radiation pattern of a single unit element groupwith m = 3 is shown in Fig. 9. The radiation patternof the unit element group is similar to that of a single
microstrip antenna. This is because element spacingd is sufficiently small. A beam tilt to the right isobtained when the bias voltage is low. Figure 10

shows the peak direction of the radiation pattern versus the bias voltage. The tilt direction is the same as
the beam scanning direction of the array with several
unit element groups, and this hardly affects the beamscanning performance of the MUSCAT array.
3.2 Measurement of scattering parameter inunit element group
To confirm the validity of the proposed configuration, we measured the scattering parameters of a fabricated unit element group with m = 3. For comparison, we calculated the properties of a unit element
group with m = 1 and the conventional phase shifter

Bias voltage = 30.0 V

17.5 V0–2–4|Eφ | (dB)

ηe =

–6

14.5 V–8
12.5 V

–10
–12
90
60
φ = 180 deg.

30

0

30

60

θ (deg.)90
φ = 0 deg.

Fig. 9. Calculated radiation pattern of unit element group(zx-plane).
φ = 0 deg.
30

80
25
Peak direction (deg.)7060

ηe (%)

50
40
30
20

20
15
10
5
0

10
5

0

1

2

3

4

5

Number of elements, m

Fig. 8. Average radiation efficiency versus number of
elements m.

28

6

14
φ = 180 deg.

16

18

20

22

24

26

28

30

Voltage (V)Fig. 10. Peak direction of unit element versus group bias
voltage.

NTT Technical Review

Selected Papers[3] shown in Fig. 11(a). The conventional phaseshifter comprised a microstrip line with a low characteristic impedance of less than 20 Ω and a pair of varactor diodes located between the microstrip line andthe ground plane. To alleviate the reflection, the distance between the varactor diodes is set to λ g /4. ThisTwo varactor diodes

Port #2

MSL

λg /4

Port #1(a)
Proposed (m = 1, calculation)
0
Conventional(calculation)–5

Proposed(m = 3, calculation)|S11| (dB)

–10
–15
–20
–25
–30

Proposed(m = 3, measurement)

–35
–40

0

–50

–100

–150

Phase shift Φ (deg.)(b)phase shifter is controlled by the DC-voltage applied
to the microstrip line.
Figures 11(b) and (c) show the scattering parameters versus Φ . The results of measurements andnumerical analysis for m = 3 indicate that the unit element group can provide a phase shift over 140° withreflection |S11| of less than –15 dB (Fig. 11(b)). Figure 11(c) indicates that the phase shift |Φmax|, whichyielded a 1-dB change in ∆|S21|, is 120°. Although thechange in varactor diode reactance causes a resonant
frequency shift in the unit element group, the effecton beam scanning performance is estimated to besmall because the bandwidth of unit element group,
at which |S11| is less than –15 dB, is more than 10%.
The results in Fig. 11(b) indicate that this band is sufficiently wide for a 140° phase shift.
On the other hand, the conventional configuration
causes significant changes in |S11| and |S21| and canchange the phase by only 40° for |S11| < –15 dB. Thisis because the varactor diodes are tuned over a wide
DC-bias range and cause a loss in the low DC-voltage
range. The reason | S 11| changes is that the largechange in the varactor diode impedance causes a serious reflection in the microstrip line. The results forthe non-multi-stage configuration (m = 1) are similar
to those of the conventional configuration, indicatinga large reflection power and large attenuation in thetransmitted power when Φ changes over a wide
range. A minor improvement is observed comparedwith the conventional configuration. This is because
the current in the varactor diodes is reduced in the m= 1 configuration because the diodes are attached tothe antenna at positions where the current density isrelatively small.

Proposed (m = 3, calculation)

4. Verification of beam-scanning performance of
MUSCAT array0–1∆|S21| (dB)–2

Proposed(m = 3, measurement)

–3
–4
–5

Proposed(m = 1,
calculation)

Conventional(calculation)

–6
–7
–8

0

–50

–100

–150

Phase shift Φ (deg.)(c)Fig. 11. |S11| and ∆|S21| versus phase shift Φ in unit
element group.

Vol. 3 No. 9 Sep. 2005

This section presents the results of numericalanalysis and experiments on an array having eight
unit element groups to clarify the beam scanning performance of the MUSCAT array. Figure 12 is a photograph of the fabricated MUSCAT array. The parameters of the unit element groups are the same as
those in the previous section. The spacing of the units
is set to be half the wavelength in vacuum and a termination load is attached to the end unit through theDC-block capacitor.
The calculated and measured radiation patterns of
the MUSCAT array are shown in Figs. 13(a) and (b).
The calculated and measured values agreed except
for the beamwidth even when the beam was steered.
29

Selected Papers

60 cm

Feed port

Termination
load

Fig. 12. Fabricated antenna.
17.5 V0

14.5 V0
12.5 V–10
Bias voltage= 30.0 V|S11| (dB)|Eφ| (dB)

Measurement
Calculation–5–10

Bias voltage= 30 V

–5

–15

–15
–20

–20

–25

–25

–30

–30

60
φ = 180 deg.

300
θ (deg.)30

19 V

16.2 V

–35
0.98

60
φ = 0 deg.

0.99

1.01

1.02

f /f c(a)17.5 V1(a)14.5 V12.5 V

0–5
Measurement–5
Bias voltage= 30.0 V

–15
–20

–15

–25
–30

60
φ = 180 deg.

Bias voltage= 30 V–10|S21| (dB)|Eφ | (dB)–10Calculation

19 V
30

0

θ (deg.)3060
φ = 0 deg.

(b)

–20
0.98

0.99

16.2 V

1

1.01

1.02

f /f c(b)

Fig. 13. Radiation pattern of MUSCAT array versus bias
voltage: (a) calculated and (b) measured.

Fig. 14. |S11| and |S21| of MUSCAT array.

We did not find any large sidelobes or a mainlobesplit, which means that the traveling wave reflection
at each unit element group is sufficiently small.
The |S11| and |S21| for the MUSCAT array are shownin Fig. 14. Here, fc is the center frequency, which is

2.4 GHz. We observed that |S11| is less than –15 dBover 4% of the frequency range ( f/fc range from 0.98
to 1.02) except for 30 V. The measured |S21| is lower
than the calculated value. From this result, we canunderstand why the measured beamwidth is wider

30

NTT Technical Review

Selected Papers

Vol. 3 No. 9 Sep. 2005

20
Measurement

Actual gain (dBi)15

Calculation

27°

3 dB

10

m=3

3 dB

5

13°
0

m=1

–5
0

10

20

30

40

50

60

70

Beam direction (deg.)

Fig. 15. Actual gain versus mainlobe direction.

15∆θ (deg.)516.2 V

Calculation

10

Bias voltage = 30 V

0
–5
–10
19 V
–15
0.98

0.99

1
f /f c

1.01

1.02

(a)

4
Relative gain (dB)than the calculated beamwidth: it is because there is asteeper current-density distribution taper in the arrayin the measurement.
The actual gain versus the beam direction is shownin Fig. 15. The measured values agree well with thecalculated ones and the range of the measured beamscanning angle within a 3-dB decrease of the actual
gain is approximately 27°. The measured beam direction shifts to the right because the reactance of thevaractor diode has additional inductance due to the
conductor line in the through hole shown in Fig. 2.
The measured gain is almost the same as the calculated gain even for the difference in |S21| values between
measurement and calculation shown in Fig. 14. From
this result, we conjectured that the difference in |S21|
is caused not by the loss, but by the radiation at theunit element group. On the other hand, the beamsteering range of the non-multi-stage configuration(m = 1) is only 13°, and there is a serious gain reduction in the scanning range above 13°. When the same
beam scanning angle is required for m = 1 and m = 3
configurations, the necessary phase shift for each element in the m = 1 configuration is much larger thanthat in the m = 3 configuration, and the lossy part of
the tunable range in the varactor diode must be usedin the m = 1 configuration. Moreover, the actual gain
for m = 1 is always much less than that for m = 3
because the number of the antenna elements is fewer
than that of m = 3 and the power consumed at the termination of m = 1 is large.
Since the proposed antenna has a symmetricalstructure on the two sides of the center, the double
range of the beam scanning can be obtained byexchanging the feed port and termination port.
Figure 16 shows the frequency dependence of thebeam direction in the zx-plane and the gain in the
MUSCAT array antenna. Here, both of these properties are normalized by fc. The ∆θ in Fig. 16(a) is thebeam direction error from the beam direction at fc. A
negative ∆θ value means that the beam direction isshifted in the direction of φ = 180°, and a positive ∆θvalue means it is shifted toward φ = 0. We found thata beam direction error ∆θ of ±11° occurred in therange of 4% bandwidth. Less than ±5° is seen in therange of 2% bandwidth. Because the width of themainlobe is about 15° according to the radiation pattern in Fig. 13, the use of a 2% bandwidth for the fabricated MUSCAT array with eight unit element
groups is adequate. If more bandwidth is required,fewer unit element groups should be used. Figure16(b) shows the gain difference for the MUSCAT
array: the gain difference within ±3 dB is shown overCalculation
16.2 V2

0–2Bias voltage = 30 V

19 V

–4
0.98

0.99

1
f /f c

1.01

1.02

(b)

Fig. 16. Frequency dependence of radiation pattern:(a) beam direction error ∆θ and (b) gain difference.

31Selected Papers

the range of a 4% bandwidth and that within only ±1
dB is shown over the range of a 2% bandwidth.
The above results indicate that the proposed MUSCAT array can achieve easy operation and a wide
beam scanning range with a simple geometry.

[6]
[7]

uration,” 2002 IEEE Int. Symp. Antennas and Propagat. Soc. i-02, pp.
251-254, Nov. 2002.
http://www.zeland.com/
I. J. Bahl, “Build microstrip antennas with paper-thin dimensions,”
Microwaves, Vol. 18, pp. 50-63, Oct. 1979.

5. Conclusion
This paper proposed a MUSCAT array using unit
element groups comprising multi-stage microstrip
antennas with tunable reactance devices that achieves
a wide beam scanning range with low loss and easy
operation. We verified the performance of the unit
element group, which consists of several microstrip
antennas and is the basic element of the MUSCAT
array. Numerical analysis of the unit element group
showed that the multi-stage configuration of the unit
element group extends the phase shift range to 120°,
which is three times that of the conventional configuration in the range of 1-dB |S21| variation and yields
more than 50% radiation efficiency for a configuration with three or more stages. We also found that theradiation pattern of the multi-stage configuration isalmost the same as that of a single microstrip antenna.
We measured the radiation pattern of a fabricated
MUSCAT array with eight unit element groups. Thisantenna achieved a wide range beam scanning of 27°,
which is more than twice the beam scanning range ofthe non-multi-stage configuration. Since this antenna
has a low-loss configuration, the discrete phaseshifters with high insertion loss, which usuallyrequire low-noise amplifiers, are not necessary; thatis, no complex bias control circuit is required. This
design yields a simple and low-loss planar phasedarray antenna that can be a key component of a lowprofile satellite tracking system.

Naoki Honma
NTT Network Innovation Laboratories.
He received the B.E., M.E., and Ph.D. degrees
in electrical engineering from Tohoku University, Sendai, Miyagi in 1996, 1998, and 2005,
respectively. He joined NTT Radio Communication Systems Laboratories in 1998. He received
the Young Engineers Award from the Institute ofElectronics, Information and Communication
Engineers (IEICE) of Japan in 2003 and the
APMC Best Paper Award in 2003. His current
research interest is planar antennas for highspeed wireless communication systems. He is amember of IEEE and IEICE.

Tomohiro Seki
Research Engineer, Wireless Systems Innovation Laboratory, NTT Network Innovation Laboratories.
He received the B.E. and M.E. degrees in electrical engineering from Tokyo University of Science, Tokyo in 1991 and 1993, respectively. In
1993, he joined NTT Radio Communication Systems Laboratories (now NTT Network Innovation Laboratories), Kanagawa. He has been
engaged in the development of array antennas
and active antennas for the millimeter-wave andmicrowave bands. He received the Young Engineer Award from IEICE in 1999. He is a memberof IEEE and IEICE.

References
Koichi Tsunekawa
[1]

[2]

[3]

[4]

[5]

32

W. H. Jones and M. L. Chapelle, “Connexion by BoeingSM-broadband
satellite communication system for mobile platforms,” IEEE MilitaryCommunications Conference (MILCOM 2001.), Vol. 2, pp. 755-758,
Oct. 2001.
F. Nagase, H. Tanaka, M. Nakayama, T. Seki, H. Kazama, and H.
Mizuno, “Mobile multimedia satellite communication system,”
IEICE Trans. Commun., Vol. E84-B, No. 4, pp. 903-909, Apr. 2001.
J. R. James, G. D. Evans, and A. Fray, “Beam scanning microstrip
arrays using diodes,” IEE Proc. H, Vol. 140, No. 1, pp. 43-51, Feb.
1993.
N. Honma, F. Kira, T. Maruyama, K. Cho, and H. Mizuno, “Compact
six-sector antenna employing patch Yagi-Uda array with common
director,” 2002 IEEE Antenna and Propagation Society International
Symposium, Vol. 1, pp. 26-29, June 2002.
N. Honma, F. Kira, T. Maruyama, and K. Cho, “Compact planar foursector antenna comprising patch Yagi-Uda arrays in a square config-

Senior Manager, Executive Research Engineer, Wireless Systems Innovation Laboratory,
NTT Network Innovation Laboratories.
He received the B.S., M.S., and Ph.D. degrees
in engineering science from the University ofTsukuba, Tsukuba, Ibaraki in 1981, 1983, and1992, respectively. He joined the ElectricalCommunications Laboratories of Nippon Telegraph and Telephone Public Corporation (nowNTT), Tokyo in 1983. Since 1984, he has been
engaged in R&D of portable telephone antennas
in land mobile communication systems. From1993 to 2003, he was with NTT DoCoMo, Inc.
where he worked on radio propagation research,
intelligent antenna systems for wireless communications, and the development of IMT-2000
antenna systems. His research interests are antenna systems for MIMO transmission, millimeterwave broadband access, and ubiquitous wirelesssystem. He is a member of IEEE and IEICE.

NTT Technical Review