Loss Analysis of T-type NPC Inverter with Active
Power Decoupling Capability Operated in
Discontinuous Current Mode
Akiyoshi Omomo, Hoai Nam Le, Keisuke Kusaka and Jun-ichi Itoh
Dept. of Electrical, Electronics and Information Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japan
itoh@vos.nagaokaut.ac.jp

Abstract—This paper proposes an active power decoupling(APD) method based on discontinuous current mode (DCM) in
T-type NPC inverter. The long lifetime is expected because
electrolytic capacitors are not required in the system. Inaddition, the APD control is achieved without any additionalinductors. The operation of the APD control is confirmed by theexperiment with a prototype. The 100-Hz component of theinput current is reduced by 89.1% compared to the result
without APD control whereas THD of the output current is1.81%. Finally, the power loss of the system is theoreticallycalculated with the calculation and the simulation. The
calculated power loss shows the good agreement with themeasured loss with an error rate of less than 10%.
Keywords—DC micro grid, discontinuous current mode,
active power decoupling, T-type NPC inverter

I. INTRODUCTION
Nowadays, DC micro grid systems have been actively
studied for the effective use of renewable resources andefficient conversion [1–7]. In the initial stage of the DC grid
employment, the use of conventional households appliances isexpected. However, conventional appliances can be connected
to only AC input. Thus, single-phase inverters are required
between DC grids and the appliances. In single-phase
inverters, the power ripple occurs in their DC-link stage dueto the difference in instantaneous power between the input andoutput power. In order to suppress the power ripple, bulky
electrolytic capacitors are generally used in the DC-link stage.
However, the employment of the electrolytic capacitors
shortens the system lifetime [8–9].
In order to eliminate electrolytic capacitors, the active
power decoupling (APD) method which uses only small
capacitance, is proposed [10–13]. The system lifetime
becomes longer by the employment of long lifetime capacitorse.g., film or ceramic capacitors, instead of electrolyticcapacitors. However, the additional circuit is required tocontrol the buffer capacitor voltage. In particular, theadditional inductors in the additional circuit restricts thesystem minimization and efficiency improvement.
So far, the APD method based on discontinuous current
mode (DCM) for a T-type NPC inverter has been proposed[14]. As the feature of the method, the additional inductors are
not required to control the buffer capacitor voltage. The output
current and neutral point current which achieves APD control,
are simultaneously controlled by using the time-sharing
technology of DCM. In [14], the operation of the proposed
method has been confirmed with the experimental result.
In this paper, the system loss characteristics are analyzed
to contribute to the optimal circuit design depends on thecircuit parameters. In the proposed method, the current

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flowing to the inverter side and the current flowing to thecapacitors, are changed when the DC-capacitors are changed.
The power losses on the switching devices and the capacitors
regarding the capacitance for APD is analyzed.
II. OPERATION PRINCIPLE
In this section, the simultaneously operation method toachieve output current control and APD control is explained.
The output power is expressed as

pout =

Vout I out(1 − cos2ωt )2(1),where Vout and Iout are the amplitudes of the output voltage andthe current, respectively, and ω is the angular grid frequency.
In the case of the single-phase inverters, the output powerfluctuates whereas the input power is constant.
Figure 1 shows the principle of the APD method. Thepower ripple is suppressed by controlling the active buffer
power that equals to the difference between the input andoutput power.
Figure 2 shows the circuit configuration of the T-type NPC
inverter. The APD control is achieved by oscillating twocapacitor voltages in opposite phases. The two capacitor
voltages VC1 and VC2 are expressed as,

VC1 =

Vdc
+ Vm sin(ωt + δ )2(2),VC 2 =Vdc− Vm sin(ωt + δ )2(3),where Vdc is the DC-link voltage, Vm is the amplitude of thecapacitor voltages, δ is the initial phase difference. Thecapacitor voltages are fluctuated by controlling the capacitor
current. The capacitor currents are calculated by,

iC1 = C

dVC1= CωVm cos(ωt + δ )
dt

(4),

iC 2 = C

dVC 2= −CωVm cos(ωt + δ )
dt

(5).

As shown in the above capacitor current equations, thecapacitor voltage control is achieved by controlling theneutral-point current in. In addition, according to (2)-(5), theactive buffer power is expressed as
pC = CωVm 2 sin(ωt + δ ) cos(ωt + δ )1= CωVm 2 sin(2ωt + 2δ )2(6).

According to (6), pC must be the same as the fluctuation
component in (1). In (6) and (1), the variable is the sinusoidal
function term. Thus, the constant value and the variable arerespectively compared. The constant value, which indicatesthe amplitude, and the fluctuation values, which indicates thephase, are respectively compared in (7)-(8),− cos(2ωt ) = sin(2ωt + 2δ )(7),

Vout I out = CωVm 2(8).

According to (7), the initial phase difference of the neutral
point current δ is calculated. In constant, the amplitude of thecapacitor voltage Vm is determined according to (8). Then, Vmis expressed asVm =Vout I outCωpout =Pin =Vm I m2

Vm I m(1 − cos 2ωoutt )2Fig. 1. Principle of power ripple compensation in single-phase DCAC converter.

(9).

Note that Vm must not exceed the half of the DC-link voltage
Vdc because the DC-link voltage is the sum of the twocapacitor voltages.
According to (4)-(5) and (9), the neutral point current whichflows to the neutral point is calculated asπin = 2 Vout I out Cω cos  ωt − 4Fig. 2. Single phase T-type NPC inverter. The output current and neutral
point current are controlled by only inducotors L1 and L2(10).

Figure 3 shows the inductor current waveform in one
switching period. In case that APD method is employed ininverters, the additional inductors are generally required tocontrol the current of the active buffer. In contrast, theadditional inductors are not required in the T-type NPC
inverters. The output current and the neutral point current
which achieves the APD control, are simultaneouslycontrolled by the output filter inductor. As shown in Fig. 3, theoutput current is controlled in a very short time in a switchingperiod. The zero current period appears after output currentcontrol. The other current which is the neutral point current iscontrolled in the remained zero current period. Therefore, twocurrent controls do not interfere with each other.

Fig. 3. Waveforms of inductor current. The output current and theneutral point current are simultaneously controlled by one inductor.
TABLE I.

SYSTEM PARAMETERS

III. EXPERIMENTAL RESULT
Table I shows the experimental parameters. Generally, thepercentage impedance %ZL of the grid-tied inductor isdesigned around 5%. However, as shown in Table I, the output
filter inductor of the prototype is 45 μH (the series connectionof two 22.5-μH inductors). Therefore, %ZL is 0.16%.
Furthermore, the additional inductor which is required forconventional power decoupling control is unnecessary.
Figures. 4 (a) and (b) show the waveforms of the output
voltage, the output current, inductor current, and the input
current with and without the proposed power decoupling
method at a 1-kW load, respectively. The output voltage iscontrolled resulting in the sinusoidal waveform at 50 Hz
whereas the fluctuation of the input current is compensated by
the proposed method. Total harmonics distortion (THD) of theoutput current is 1.81%.
Figure 5 shows the harmonic analysis of the input current.
According to Fig. 5, by using the proposed method, thefluctuation component of 100 Hz, i.e. twice the grid frequency,
is reduced by 89.1% compared to that without the APD
method. In addition, any other harmonic components do not
exceed 10%.

IV. LOSS ANALYSIS
In this section, the loss analysis of the system losses isintroduced. The losses in the system can be divided into theconduction losses, the switching losses, and the inductor
losses.
The conduction losses of the on-resistance in the switching
devices and the ESR of capacitors are calculated by
Pcond = RI RMS 2(11).

The current which flows to the devices and the capacitors
are same as a part of the inductor current. The DCM inductor
current is expressed as
4 f sw / f grid

I L _ RMS = f grid   
m=1

n=1

Tswdm

02 Vn  t  dt
L (12),(a) Operation without power decoupling control.
(b) Operation with power decoupling control.
Fig. 4. Waveforms of input current, output voltage, output current and inductor current in steady state.

Fig.5 Harmonic components of input current

where fgrid is the grid frequency, fsw is the switching frequency,
Tsw is the switching period, d is the duty of device on time, Vn
is the potential difference between both ends of inductors andL is the filter inductor. The summation of the integral terms isthe same as the RMS current of the devices and DC-link
capacitors, which are determined by the switching patterns.
For instance, when the polarity of the output current ispositive, the current flows to SW1 in the term during d1,d2,Hence, the RMS current of SW1 is calculated as

I SW 1_ RMS =f grid 2 fsw / f grid Tswdm  Vn 2
 t  dt
2 m=1 n=1 0  L (13).

The other switches and capacitor’s RMS current arecalculated by the common manner.
The calculation of the switching losses based on DCM isdifficult due to its nonlinearity. Thus, in this paper, the circuit
simulation software “PLECS” is used to calculate theswitching losses. The turn-on and turn-off losses are extracted
from a datasheet of the devices that is used in the prototype.
Similarly, the inductor losses in DCM is obtained by theanalysis software “Gecko-simulations”. The specification ofthe inductor is based on the inductor which is used in theprototype.
V. LOSS COMPARISON AND CAPACITANCE DESIGN
A. Loss comparison
Figure 6 shows the loss comparison result between thecalculation and the experimental result. As shown in Fig. 6,
the error rate between the calculated loss and the measured
loss is less than 10%. Furthermore, it can be observed that theconduction losses of the switching devices and the inductor

copper loss dominates a large proportion of the system loss.
There, these losses should be focused to be reduced to further
improve the efficiency of the system.
B. Capacitance design for highest efficiency
According to (9)-(10), the current flowing to the inverter
side iinv is expressed asiinv = Vout Iout VDC − ωCVout Iout cos(ωt + δ )(14).

According to (11), the conduction loss increases when the DClink capacitor C is increased because the amplitude of iinv, iC1,and iC2 are increased. However, the capacitor ESR loss isreduced by paralleling capacitors because the ESR of thecapacitors is reduced. In this paper, the relationship between
the device loss and the capacitor ESR loss is clarified.
Figure 7 shows the characteristics of the total loss versus theDC-link capacitance. According to (2)-(3), Vm must notsurpass the half of DC-link voltage VDC/2. According to (9),Vm is in inverse proportion to C0.5 when the rated power isdecided. In this paper, the lower limit of the capacitance isdesigned as 100 μF that Vm is lower than 90% of VDC/2.
The horizontal axis of Fig. 7 shows the parallel number ofthe capacitors and the capacitance simultaneously. In Fig. 7,
the conduction loss dominates the system loss in any cases.
According to (14), the amplitudes of iinv, iC1, and iC2 change
by changing the capacitance. Thus, the loss reduction isexpected by reducing the capacitance. Meanwhile the effect of
the capacitance variation on the switching loss is small
because the charging and discharging energy of the capacitor
is constant. Note that the capacitor loss causes due to its ESR.

The capacitor ESR loss is reduced by paralleling thecapacitors due to the reduction of ESR. However, according
to (10) and (14), the reduction of the capacitor ESR loss issmall because the amplitudes of iC1, and iC2 are increased byparalleling capacitors.
Consequently, the relation between the conduction loss
and the capacitor loss is a trade-off relation. However, thevariation of the conduction loss is dominant in total losses.
Therefore, it can be concluded that a small capacitor for theDC-capacitor will decrease the total power loss according to(9) and (14).
VI. CONCLUSION
This paper presented the APD method for the T-type NPC
inverter operated in DCM. The achievement of the APD
control without additional magnetic components has been
confirmed in the experiment. The calculated system loss iscompared to the measured loss. The error rate between thecalculated loss and the measured loss is less than 10%. Inaddition, the total loss of switching devices versus thecapacitance of the DC-link capacitors, was analyzed. As aresult, the total loss of switching devices was reduced inaccordance with the capacitance reduction of the DC-link
capacitors. In future work, the efficiency will be furtherimproved.

Fig. 6. Loss comparison between experiment and calculation. The error
rate between Experiment and calculation is less than 10%.

REFERENCES
[1]

[2]

[3]

[4]

[5]

[6]

[7]

[8]

[9]

Y. Ito, Z.Yang, and H. Akagi, “DC micro-grid based distribution powergeneration system”, IEEE Int. Power Electron. Motion Control Conf.,
Vol. 3, pp. 1740–1745 (2004)
X. Lu, J. M. Guerrero, K. Sun, “An Improved Droop Control Method
for DC Microgrids Based on Low Bandwidth Communication With
DC Bus Voltage Restoration and Enhanced Current Sharing
Accuracy”, IEEE. Trans Vol. 29, pp. 1800-1812 (2014)
T. Dragičević, J. M. Guerrero, J. C. Vasquez, “Supervisory Control ofan Adaptive-Droop Regulated DC Microgrid With Battery
Management Capability”, IEEE. Trans, Vol. 29, pp. 695-706 (2014)
X. Lu, K sun, J. M. Guerrero, J. C. Vasquez, “State-of-Charge Balance
Using Adaptive Droop Control for Distributed Energy Storage Systems
in DC Microgrid Applications”, IEEE. Trans, Vol. 61, pp. 2804-2815(2014)
V. Nasirian, S. Moayedi, A. Davoudi, F. L. Lewis, “Distributed
Cooperative Control of DC Microgrids”, IEEE. Trans Vol. 30, pp.
2288-2303 (2015)
Toru Tanaka, Yoshinori Takahashi, Kenji Natori, and Yukihiko
Sato”High-Efficiency Floating Bidirectional Power Flow Controller
for Next-Generation DC Power Network,” IEEJ J. Industry
Applications, vol. 7, no. 1, pp. 29-34, (2018)
Toshiki Nakanishi, and Jun-ichi Itoh,”Control Strategy for Modular
Multilevel Converter based on Single-phase Power Factor Correction
Converter”,IEEJ J. Industry Applications, vol.6, no.1, pp.46-57, (2017)
H. Hu, S. Harb, X. Fang, D. Zhang, Q. Zhang : "A Three-port Flyback
for PV Microinverter Applications With Power Pulsation Decoupling
Capability", IEEE. Trans, Vol. 27, No. 9, pp. 3953-3964 (2012)
Kodai Abe, Hitoshi Haga, Kiyoshi Ohishi, and Yuki
Yokokura,”Current Ripple Suppression Control Based on Prediction ofFig. 7. Loss analysis result. The total loss is reduced in accordancewith minimizing C.

Resonance Cancellation Voltage for Electrolytic-Capacitor-LessInverter”,IEEJ J. Industry Applications, vol.6, no.1, pp.1-11, (2017)[10] Y. Tang, F. Blaabjerg, P. C. Loh, C. Jin, P. Wang, “Decoupling ofFluctuating Power in Single-Phase Systems Through a Symmetrical
Half-Bridge Circuit”, IEEE. Trans, Vol. 30, No. 4, pp.1855-1865(2015)[11] H. Hu, S. Harb, N. Kutkut, I. Batarseh, Z. J. Shen : “Power DecouplingTechniques for Micro-inverters in PV Systems — a Review,” Energy
Conversion Congress and Exposition 2010, pp.3235-3240, pp. 12-16(2010)[12] K. H. Chao, P. T. Cheng: "Power Decoupling Methods for single-phase
three-poles AC/DC converters", Energy Conversion Congress andExposition 2009, pp. 3742-3747 (2009)[13] S.Yamaguchi, T.Shimizu: “Single-phase Power Conditioner with a
Buck-boost-type Power Decoupling Circuit”, IEEJ Journal of Industry
Applications, Vol. 5, No. 3, pp. 191-198 (2016)[14] A. Omomo, N. Takaoka, H. N. Le, K. Kusaka, J. Itoh, “T-type NPC
Inverter with Active Power Decoupling Capability Using
Discontinuous Current Mode”, IEEE EPE2018, pp. 1-10 (2018)