DC/DC Boost Converter Functionality in a
Three-phase Indirect Matrix Converter
Goh Teck Chiang* and Jun-ichi Itoh**Nagaoka University of Technology, Niigata, Japan
Abstract–
An indirect matrix converter connected with two input power sources is proposed; a gasoline generator as
the main AC power supply and batteries as the secondary power source. The indirect matrix converter issmall in size because of having a DC link part without an electrolytic capacitor. The DC link part isutilized by connection with a boost up chopper with batteries as a secondary input power source.
Furthermore, the chopper connects to the neutral point of the motor and utilizes the leakage inductance ofthe motor as a reactor component. The proposed technique successfully further reduce the size of theconverter by removing the boost reactor in the boost converter stage. The proposed converter is simulated
and experimentally validated using a 750 W prototype and an induction motor driven with V/f control.
The total harmonic distortion (THD) of the input and output currents are 4 and 3.7%, respectively, and theefficiency is 96%.

Index TermThree-phase AC/AC converter, Neutral point of a motor, Zero vector switching, Boost-up chopper,
Leakage inductance*Nagaoka University of Technology, Nagaoka, Japan
Telephone: +81-258-47-9563
FAX: +81-258-47-9563
E-mail: tcgoh@stn.nagaokaut.ac.jp (Goh Teck Chiang)
itoh@vos.nagaokaut.ac.jp (Jun-ichi Itoh)
This paper was presented at ECCE 2009, San Jose, U.S.A, and modified the content for the IEEE
transaction submission

INTRODUCTION
Environmental responsibility has become a significant concern for communities, so that thedevelopment of renewable power sources, such as wind turbines and low-carbon-emission hybrid electricvehicles (HEVs) is progressing rapidly. One of the most common applied converters in hybrid systems is
the AC/DC/AC converter, because it has the ability of connection to two different power sources. The
generator mainly supplies constant power to the load and a battery is used as an alternate power source todrive an electric motor and also to absorb the power fluctuation during periods of high peak energy
demand.
Figure 1 shows a conventional AC/DC/AC power converter, which typically consists of a pulse widthmodulation (PWM) rectifier, a DC link capacitor and a PWM inverter, also known as a back-to-back(BTB) system [1-3]. The PWM rectifier is often used to reduce the harmonic currents in a generator andcontrol the DC link voltage [4-7]. In order to obtain high performance under an adjustable speed drive
system, a constant DC link voltage is required in a BTB system, because the voltage fluctuation of the DC
link part will cause an output voltage error. A typical method for reduction of the voltage fluctuation is toplace a large electrolytic capacitor into the DC link part as a filtering device between the rectifier and theinverter. However, a large electrolytic capacitor is bulky.
Another approach is to reduce the capacity of the electrolytic capacitor by application of a high speed
DC voltage controller to the rectifier control [8]. However, the control response is limited by the delay ofthe voltage detection and digital controller; therefore, the electrolytic capacitor is still required. In
addition, the capacitance is not reduced since the DC link capacitor is dominated by the capacitor current.
As a result, a large amount of space is required for the capacitor installation in a practical device. In
addition, electrolytic capacitors are not suitable for high temperature applications, such as in HEVs.
Overall, these disadvantages of the electrolytic capacitor affect the reliability of the converter.
For the secondary input power source, a boost converter which consists of a boost reactor and aswitching leg (IGBT) is connected with batteries to the DC link part of the BTB system. Boost converter
will control the battery current and the battery power will be used as a secondary power to drive theelectric motor.
In this paper, a new circuit topology is presented, which is composed of an indirect matrix converter
and a DC/DC boost converter that connects to the neutral point of a motor. An indirect matrix converter(IMC) has high efficiency and is easily configured in comparison to matrix converters [9-14]. In addition,
this converter does not require a DC link electrolytic capacitor to filter the DC ripple voltage. It uses a
direct conversion technique where the frequency of the DC link voltage contains a ripple with six times of
the input frequency. However, the output voltage transfer ratio is limited by this direct conversion
technique which is similar to the matrix converter, where output voltage = 0.866 of the input voltage [15].

Nevertheless, an appropriate control over the inverter is also proposed, so that it is possible to connect
a DC chopper to the neutral point of the motor and to operate as a DC/DC converter [16]. This DC/DC
converter with a battery is performed as a secondary power source of the IMC to drive the electric motor.
The proposed circuit utilizes the neutral point of a motor in the boost converter, because the leakage
inductance of the motor can be used as a reactor. Generally, the leakage inductance is around 10% of therating impedance in an induction motor. For the proposed DC converter, around 3% of the reactor isenough to use as a boost reactor component. Please note that the synchronous reactance in a permanent
motor is higher than the leakage inductance of an induction motor.
By removing the electrolytic capacitor and the boost-up reactor, the remaining part of the proposed
circuit is constructed only of silicon components, namely insulated gate bipolar transistors (IGBTs) anddiodes. As a result, the proposed circuit is high efficiency and high reliability. Simulation andexperimental results clearly demonstrate that the circuit is capable of providing sinusoidal waveforms forthe input and output, and high efficiency and a high power factor can be achieved.
PROPOSED CIRCUIT TOPOLOGY
Figure 2 shows the proposed circuit configuration. The indirect matrix converter can be simply divided
into primary and secondary stages. The primary stage for the AC power source consists of twelve units ofreverse blocking IGBTs [17], also known as a current source rectifier, where bidirectional power flow ispossible in this circuit structure. A LC filter is required at the input of the primary stage to smooth the
input current. The secondary stage for the motor consists of six IGBT units, which is similar to a standard
voltage source inverter. The advantage of this converter over a BTB is that the primary side does notcontain switching loss because zero current switching can be applied. The switching timing of the
primary side is during the zero-current period of the DC link when the secondary stage output zerovoltage. Therefore, high efficiency is achievable in this converter [12].
The other reason to use the indirect matrix converter is that the indirect matrix converter has a DC linkpart which is difference to the conventional matrix converter. The DC link part is utilized by adding a

Fig. 1. Back-to-back converter.

Fig. 2. Proposed circuit topology.

boost converter to the indirect matrix converter. The boost converter connects to the battery and the otherterminal of the battery is then connected to the neutral point of the motor.
A snubber circuit is also included in the DC link part to absorb the voltage overshoot from reactive
elements in the circuit [18]. It is used to prevent damage to the switching devices in the secondary side
due to a sudden large voltage. It should be noted that the capacity of the snubber capacitor is smaller thanthe DC link capacitor in a BTB system, because the ripple current of the DC link part does not flow in thesnubber capacitor.
The chopper circuit is connected in the DC link and batteries are connected to the neutral point of the
motor. The leakage inductance of the motor is used as a boost up reactor in the proposed circuit. As aresult, the proposed converter does not require bulky passive components.
CONTROL STRATEGY
Figure 3(a) shows a control block diagram of the proposed circuit. The primary side, the DC chopper
and the secondary side are individually controlled by their own commands. A carrier comparison method
is used as the PWM modulation, according to the control strategy [19]. The relationship between the
output and input voltages is obtained by (1). The secondary side operates as a four-phase voltage source
inverter by addition of the DC chopper as the fourth leg,
⎡ vu ⎤ ⎡ sup⎢ v ⎥ ⎢s⎢ v ⎥ = ⎢ vp
⎢ vw ⎥ ⎢ swp⎢ ⎥ ⎢⎣vbat ⎦ ⎢⎣ scp

sun ⎤svn ⎥⎥ ⎡ srp
swn ⎥ ⎢⎣ srn⎥
scn ⎥⎦

ssp
ssn

⎡vr ⎤
stp ⎤ ⎢ ⎥vsstn ⎥⎦ ⎢ ⎥⎢⎣ vt ⎥⎦

(1)

where sxy represents the switching function of the switches. When sxy is turned on, sxy = 1 and when sxy isturned off, sxy = 0.
A. Primary side control
The primary side controller is designed with a current-type PWM rectifier command. It uses a pulse
pattern conversion to convert the PWM pulses of the voltage source type into the PWM pulse of thecurrent source type by a simple logic selector. It uses a single leg modulation where the switching period
can be reduced from 2π/3 to π/3, where the 2π/3 is the switching period of the conventional two-phasemodulation [17]. That is, the leg with the maximum input phase voltage will always be turned ON, andthe other two legs will be always turned OFF as shown in Figure 3(b). When the maximum input phasevoltage is changing, (for example, from +R-phase to -S-phase), the related max phase voltage leg and the
mid phase voltage leg will be switched at zero current until the relevant switch that contains the midphase voltage becomes the maximum input phase voltage. From this direct conversion technique, a DC
link voltage that contains a ripple with 6 times of the input frequency will be formed [17].
B. Secondary side control

Fig. 3(a). Control block diagram.

Fig. 3(b). Primary side switching pattern.

A conventional controller method for a voltage source type inverter is applied to the DC chopper andthe inverter with a lean controlled carrier modulation. The carrier modulation forms a new carrier where
the peak position of the triangular carrier is controlled by the duty ratio of the rectifier side pulse. This
rectifier pulse is used to control the switching timing of the primary stage and the zero vector of thesecondary stage. From the control, zero current switching is achieved in the primary stage where the dc
link current becomes zero at the peak of every carrier. This new carrier is then used in the secondary sideand the DC chopper side as a normal PWM comparison method, also referred to as an inverter carrier.
The boost converter is not a stand-alone circuit in the proposed circuit. Operation is strongly dependent
on the secondary side of the indirect matrix converter. Zero vector outputs on the secondary side are thekey factor to link the boost converter to the indirect matrix converter. The zero vector controls theamplitude of the output voltage. There are two functions of the zero vector output to the secondary side.
The first is to implement zero current switching on the primary side, so that switching losses do not occur
at the primary side. The second function involves operation of the boost converter, which will be
described in a later chapter.
Figure 4 shows an example of the relationship between the normal carrier applied to the primary sideand the new inverter carrier applied to the secondary side. The inverter commands are given by thevoltage controller as described in Chapter IV. It is noted that the DC chopper is controlled as the fourth
leg of the inverter, so that the DC chopper command is compared by the same carrier with the inverter
voltage commands. There are two methods to generate an inverter carrier; Figure 4(a) represents thesymmetrical type, which has approximately double the switching frequency of the rectifier switchingfrequency, and Figure 4(b) shows the asymmetrical type, which has the same switching frequency as therectifier switching frequency [18].
In Figure 4(a), the bottom peak position of the triangular carrier is controlled by the duty ratio of therectifier pulse, as shown in the upper part of the figure. The chopper commands, along with the inverter
output voltage commands, are compared with this new inverter carrier to obtain the desired switchingpatterns. The zero vector periods are shown in the lower part of Figure 4(a). The switching pulses of thesecondary side attain the zero vectors for every carrier cycle. The primary side arms switch at every zero
vector period.
In Figure 4, zv_u and zv_l represent the zero vector periods of the inverter, where zv_u =Sup=Svp=Swp=1 (upper arm zero vector) and zv_l = Sup=Svp=Swp=0 (lower arm zero vector(Sun=Svn=Swn=1)). The upper arm of the chopper (Scp) switches ON at every zero vector period of
zv_u. On the other hand, the lower arm of the chopper (Scn) will switch ON at every zero vector periodsof zv_l. During these zero vector periods, the boost converter is operated in the on-state, and the battery
current through the leakage inductance of the motor increases. During the non-zero vector periods, also
known as the Off-state operation, the battery current is released into the capacitor in the LC filter at thepower source. The operation state in the figure is referred to the boost converter operation.
When the switching frequency of the rectifier is 10 kHz, the control method applied in Figure 4(a)generates a new symmetrical carrier that has a frequency of approximately 20 kHz. This is approximately
twice the primary side switching frequency. Alternatively, according to Figure 4(b), an inverter carrier canbe formed based on the duty of the rectifier command, which is asymmetrical with a frequency of 10 kHz.
By comparing the symmetrical and asymmetrical inverter carriers in Figure 4, it should be noted thatthe zero current switching in the rectifier is not affected by the inverter carrier, because both carriers areformed following the rectifier duty. Since every carrier time is longer in the asymmetrical inverter carrier,the sequence of the zero vector periods becomes slower; therefore, the boost converter will achieve better
efficiency, but the current ripple in the battery will be increased. Further, the asymmetrical method canachieve better total harmonics distortion (THD) values for the output because the dead time effect issmaller due to the lower switching frequency.
The other disadvantage of the asymmetrical inverter carrier is the detection of the load current. Usually,
the average value of the load current appears at the peak of the symmetrical inverter carrier, so that it canbe easily detected using the symmetrical inverter carrier. However, for the asymmetrical carrier, theaverage current point does not agree with the peak of the asymmetrical carrier; therefore, in order todetect the average current, a low-pass-filter is required. Consequently, control performance will bedecreased.
UTILIZATION OF THE NEUTRAL POINT OF THE MOTOR

The voltage commands of the secondary stage are decided by the battery current command and the10 kHz
Carrier

Rec*

Srp
Ssp
Stp

Rec*

Srp
Ssp
Stp

vu_refvv_ref

20 kHz Inverter Carrier
vu_refvv_refvw_refvc_refvw_refvc_refzv_u zv_l

Scp
Sup
Svp
Swp
Operation
state
ibat

10 kHzCarrier

10 kHz Inverter Carrier

zv_u

zv_l

Scp
Sup
Svp
Swp
On On OnOn

On

Off Off Off(a). Symmetrical inverter carrier.

ibat

On
Off

On

On
Off

Off

(b). Asymmetrical inverter carrier.

Fig. 4. Relationship between the zero vectors and boost converter operation.

output voltage commands for the three-phase load. A three-phase inverter has eight output voltage space
vectors, including two zero vectors. The importance of the zero vectors explains the behavior of the boost
converter along with the neutral point of the motor. The boost converter will operate at every zero vectorof the secondary side of the converter. On the other hands, the output voltage for the three-phase load iscontrolled by other voltage vectors as shown in [16].
Figure 5 illustrates the output current flow diagram of the secondary side under the normal operation
and Figure 6 shows the zero-phase sequence equivalent circuit where the battery is operating at discharge
mode. In Figure 5, the secondary side functions as a conventional three-phase inverter with a motor; it
controls the motor speed and torque. Zero-phase sequence is happening at every zero vector periods of thesecondary side. The current at the neutral point of the motor is zero and a positive or negative battery
current can be controlled as shown in Figure 6. Note that the polarity of the battery voltage can beconnected in facing the neutral point of the motor or in a reverse way [16]. Cin represents the capacitorsfrom the LC filter; since two switches in the primary side will always be turned ON, the capacitors Cin
can be considered in the DC link voltage. On the other hand, for the zero-phase sequence, the motor line
voltage can be considered as zero, so that the motor can be considered as a leakage inductance. In
addition, the secondary side of the converter can be considered as a single leg topology. The batterycurrent first goes into the secondary side and flows out through the neutral line and charges or discharges
the battery. The battery current can be controlled by the PI controller.
The zero vectors are two particular vectors that generate zero line voltage to the motor. The neutral
point voltage v0 of the motor, based on the neutral point of the DC link part, is obtained by (2)E dc⎧⎪⎪vo = 2⎨⎪v = - E dco2⎩⎪

When all upper arms are ON(2)
When all lower arms are ON

where Edc is the DC link voltage and vo is the voltage of the neutral point of the motor, based on theneutral point of the DC link part.
A high DC link voltage is mandatory in order to control the zero vectors; therefore, the relationshipbetween the DC link voltage (Edc), the inverter line voltage (vinv) and the battery voltage (Vbat) will be
discussed. The inverter output voltage, vu, vv, and vw, with respect to the neutral point voltage of the DC
link, is expressed as
Edc⎧⎪ vu = a 2 sin ωt + vo
⎪
Edc
2π
⎪
sin(ωt −) + vo .
⎨ vv = a
2
3
⎪
Edc
4π
⎪
⎪ vw = a 2 sin(ωt − 3 ) + vo⎩

(3)

where a is the modulation index of the motor phase voltage, 0 < a < 1, vo is the neutral point voltage ofthe motor (during zero phase sequence), and ω is the inverter output angular frequency.
The inverter line voltage is then given by (u-v phase)
vuv = aπ3
Edc sin(ωt + ).
2
6

(4)

Eq. (5) shows the relationship for an inverter to obtain the maximum output line voltage Vinv (RMS) underthe maximum reference magnitude of a three phase modulation.

2

Edc > 2

3

Vinv .

(5)

The maximum line voltage between the inverter leg and chopper leg can be obtained as, (RMS)

vux =

2
3

Vinv + Vbat .

(6)

Because vux must be smaller than Edc, the inverter voltage and battery voltage are constrained by Eq. (7)

Edc >

2
3

Vinv + Vbat .

(7)

As a result, the DC link voltage of the proposed circuit must satisfy both requirements as shown by Eq.

(8), which can be referring to Figure 6.
⎧ 2
2Vinv
≥ Vbat
Vinv
When
⎪2
3
⎪ 3
.
Edc >⎨2Vinv⎪ 2< Vbat⎪ 3 Vinv + Vbat When
3
⎩

(8)

Note that in Eq. (8), the Vbat can always be neutralized with half of the Edc under the two conditions. Thatis, the Vbat must be always smaller than half of the Edc, since the Edc is always known as the 0.866 of the
input phase voltage, as shown in Eq. (9).

2
Vin 0.866 > Vbat
2

(9)Furthermore, a new expression of the secondary side current is given as follows, assuming that theleakage impedance is even during the zero-phase sequence equivalent circuit.
ibat⎧⎪iu = ia + 3
⎪⎪
ibat
.
⎨ iv = ib +3⎪
⎪i = i + ibat⎪⎩ w c 3(10)where iu, iv, and iw are the inverter current, ia, ib, and ic are the positive-phase inverter current and ibat is thebattery current.

Fig. 5. Secondary side current flow diagram (Normal operation).

Fig. 6. Zero phase sequence equivalent circuit.

SIMULATION RESULTS
Table I shows the simulation parameters for both results. The proposed circuit was simulated under two
conditions of battery discharge and charge by using a circuit simulator (PSIM, Powersim Technologies
Inc.). An ACR (Automatic current regulator) controller controls the battery current to a desired positive or
negative value. An ideal battery current ibat is purposely adjusted at a specific time of 38 ms to confirm the
proposed circuit performance. The motor model, which consists of three sets of voltage sources asback-electromotive forces and leakage inductances, is used in the simulation. The asymmetrical inverter
carrier was used in the simulation.
Figure 7 shows the battery discharge mode, with the battery current controlled from 0.5 to 2 A. The two
waveforms show the input power supply voltages vr, vs, and vt, the input currents ir, is, and it, and theoutput line voltages (vuv(LPF), vvw(LPF), and vwu(LPF)) through a low pass filter, which has a cut-off frequencyof 1 kHz, to observe the low frequency components, the output currents iu, iv, and iw, and the battery
current ibat. The results show that the THD of both the input and output currents are less than 4%. It
should be noted that at 20 ms, the input current magnitude decreases due to the increment of ibat, whichindicates that the increase of the battery power leads to a decrease in generator power.
On the other hand, Figure 8 shows the ACR (Automatic current regulator) controls the battery current
from 0.5 to -2 A. The battery is charged from a generator under this condition. The results are also showthat when in the charging mode, both the input and output currents have good sinusoidal waveforms. At
20 ms, as the ibat decreases, the input current is forced to increase, because higher power is required tocharge the battery. These two waveforms provide evidence of good power management between thegenerator and the battery.

Fig. 7. Simulation results (Battery = Discharge mode).

Fig. 8. Simulation results (Battery = Charge mode).

TABLE ISIMULATION PARAMETERS
Input voltage

200 V

DC source

100 V

Input frequency

50 Hz

Output voltage

173 V

Carrier frequency

10 kHzOutput frequency

35 Hz

Inverter frequency

10 kHzLeakage inductance

5 mH

TABLE II
EXPERIMENTAL PARAMETERS
Input voltage

200 V

Inductor

1.7 mH

Input frequency

50 Hz

Capacitor

2.5 μF

Carrier frequency

10 kHzDC source

100 V

Output frequency

30 Hz

LC Filter Cut-off frequency

2.4 kHzTABLE III
MOTOR PARAMETERS (FUJI: MLH6085M)
Motor Power

750 W

Rated current

3.6 A

Poles

4/ 50 Hz

Rated voltage

200 V

RPM

1420

Leakage inductance

4.42 mH

EXPERIMENTAL RESULTS
A 750 W prototype was built and tested using two operation modes, the same as those described in thesimulation section. Mode I is battery discharge under motoring operation and Mode II is battery charge
under motoring operation. Both conditions were verified using the parameters shown in Table II. The
controller used the asymmetrical format to generate the 10 kHz inverter carrier. Table III shows the
specification data for the motor. Note that this is conventional motor with no particular settings required
for the neutral point connection.
Fundamental Operations
Figures 9 and 10 show the Mode I and Mode II operations, respectively. In Figure 9, the DC power
supply is set to 100 V and the battery current is controlled to 2 A. Similarly, in Figure 10, the battery
current is controlled to -2 A. Good sinusoidal waveforms were achieved for the input and output currentof both operation modes. Figure 11 shows the input power factor of the discharge and charge modes. Bothmodes achieved input power factors of more than 98%, which can be considered as a unity power factor.
The input power factor decreases during the discharge mode, because the input current becomes smaller

Fig. 9. Experimental results (Mode I = Battery discharge).

Fig. 10. Experimental results (Mode II = Battery charge).

in comparison with the charge mode.
Figure 12 shows the input current THD and the output current THD for both modes. The input current
THD obtained during the charge mode is 3.6%, and that during the discharge mode is 4.0%. The lowest
output current THD obtained during the charge mode is 3.0% and that during the discharge mode is 3.7%.
Figure 13 shows the picture of the prototype. All switching units are mounted on top of a heat sink. The
top of the picture shows the primary side switching units. GDU boards are placed on top of the primaryside switching units. The picture also shows that the switching units for the secondary side and the boost
converter are based on a single module IGBT unit.
Motor Performance Analysis
Figure 14 demonstrates motor related experimental results in order to confirm the motor performancefor the proposed circuit. This figure shows the torque impact characteristic of the tested motor. The output
frequency is 30 Hz, and the step increase of torque is 100%. The battery power is set at 200 W, and the
input power is closely to 50 W when the torque is 0%. The input current ir shows a bad quality of
waveform due to the low input power since battery power is supplying the induction motor. When thetorque increases to 100%, the battery power maintains at 200W, subsequently the input power provides
the additional required power to keep the motor speed. The RPM waveform demonstrates a good controlof speed and the output current shows a corresponding sinusoidal waveform.
Efficiency and Loss Analysis
The loss analysis of the proposed circuit is now discussed, which is carried out using a circuit simulator(PSIM, Powersim Technologies Inc.) and Dynamic Link Library (DLL) files [20]. The analysis was
conducted for two categories, by application of the symmetrical and asymmetrical inverter carriers.
Figures 15, 16, 17 and 18 show the loss analysis results simulated under the parameters as shown in Table
I and Table IV shows the devices parameter. Note that in the loss analysis, the primary side is assumed touse RB-IGBT. The output power is 750 W and the input power ratio is 9:1 in Figs. 15 and 16. The input
power ratio is referring between the generator power and battery power respectively. A ratio of 9:1 meansthat the total input power supplied from the generator power is 90% and the remaining 10% is supplied
from the battery power.
Figure 15 shows details of the switching losses when the symmetrical inverter carrier is applied. Zerocurrent switching is implemented on the primary side, and therefore conduction loss occurs only in thatside. The total loss for the discharge mode is approximately 27 W, and that for the charge mode isapproximately 29 W. The analysis verifies that the converter can achieve an efficiency of 96.4%. A
conventional back to back converter may achieve an efficiency of approximately 93% [21].
Two reasons are found for the higher losses in the secondary side. The first is that the switchingfrequency in the secondary side is 20 kHz when using the symmetric method, as discussed in chapter III.

The second reason is that the inverter current contains the battery current during the zero phase sequence.
The higher switching frequency and larger battery current resulted in increased losses.
Figure 16 shows the results of the asymmetric inverter carrier loss analysis, which was simulated under
1.00

8

0.96

Discharge mode- Input current THD

7

THD [%]

Input Power Factor

0.98

9

Discharge mode- Input power factor

0.94
0.92
Charge mode- Input power factor

0.90

Discharge mode- Output current THD

6
5
4

Charge mode- Input current THD

0.88

3

0.86

2

200

300
400
500
OutputPower
Output
Power
[W]

600

700

Charge mode- Output current THD
200

300

400

500

600

700

OutputOutputPowerPower[W][W]

Fig. 11. Input power factor.

Fig. 12. Input and output current THD valuesN

150
r/min/div

888 r/min
T

50%/div

ir
1A/diviu
2A/div
2000ms/div

Fig. 13. Picture of the prototype

Fig. 14. Torque impact characteristic of the proposed circuit.
TABLE IVDEVICE PARAMETERS
Primary side

Secondary side

Chopper

1MBH30D-060

2MBI50N-060

2MBI50N-060(Fuji Electric)(Fuji Electric)(Fuji Electric)

Collector-Emitter voltage

600 V

600 V

600 V

Gate-Emitter voltage

+20 V

+20 V

+20 V

Collector current

30 A

50 A

50 A

Turn-on time (Typ.)0.16 μs

0.6 μs0.6 μsTurn-off time (Typ.)0.30 μs

0.6 μs0.6 μs

Reverse recovery time

300 ns

300 ns

300 nsDevices modelSwitching frequency = 10 kHz
Input power ratio = 9:1
Output Power = 750W

Input frequency = 50 Hz
Output frequency = 35 Hz

Recovery loss

Loss [W]20

Switching loss
(Turn-on)
Switching loss
(Turn-off)
Conduction loss

15
10(Discharge)(Charge)Primary

40
35(Discharge)(Charge) (Discharge)

5

Conduction loss

(Discharge)(Charge)(Discharge)(Charge)

Chopper

Secondary(Asymmetric method).

40
35

25
Secondary side

15

Inverter switching frequency =10 kHz
Output Power = 750W
Switching frequency =10 kHz

30

Loss [W]Loss [W]

Switching loss
(Turn-off)(Symmetric method).

30

DC Chopper

Total Loss

25
20
15
10

Primary side

5
0

10

Fig. 16. Loss analysis – Switching device losses

Total Loss

10

Recovery loss
Switching loss
(Turn-on)

Fig. 15. Loss analysis – Switching device losses

Inverter switching frequency =20 kHz
Output Power = 750W
Switching frequency =10 kHz

20

15

0 (Discharge) (Charge)
Primary(Charge)

Chopper

Secondary

Input frequency = 50 Hz
Output frequency = 35 Hz

20

5
0

Switching frequency = 10 kHz
Input power ratio = 9:1
Inverter frequency = 10 kHz

25

Loss [W]25

Secondary side
DC ChopperPrimary side

5
10:0

9:1

8:2

7:3

6:4

5:5

4:6

Input Power Ratio

3:7

2:8

1:9

0

10:0

9:1

8:2

7:3

6:4

5:5

Input Power Ratio

4:6

3:7

2:8

1:9

Fig. 17. Loss analysis – Changes in input ratio

Fig. 18. Loss analysis – Changes in input power ratio(Symmetric method).

(Asymmetric method).

the same conditions as those for Figure 15. The loss in the secondary side decreases approximately 20%
and the loss in the chopper decreases by approximately 25%. The total loss for the discharge mode is 23
W and that for the charge mode is 25 W. An efficiency of 96.9% can be achieved, which is an
improvement of approximately 0.5%.
Furthermore, in comparison the proposed circuit with the typical indirect matrix converter; provided
the energy flow is from AC to AC only, the efficiency is almost equivalent for the proposed circuit. But
for DC to AC, the motor loss will affect the efficiency, where a special type of motor with a connectableneutral point is required in order to reduce the copper loss in the motor.
Figures 17 and 18 show the calculations of loss analysis by powering with various input power ratiosfrom 10:0 to 1:9, which represent the ratio of the generator power to battery power. In this case, the losses
in chopper gradually increase as the battery power is going larger. The losses in primary side decrease
accordingly to the input power.
For the symmetric method in Figure 17, it is obvious that the loss in the DC chopper increases sharply

as the battery current increases. However, the loss in the primary side does not reach zero, but drops to a
constant 4 W from 7 W.
On the other hand, as the generator power is less than 225 W, the loss in the secondary side starts todecrease. This is because the output voltage is directly dependent on the input voltage in the indirect
matrix converter. As the input voltage decreases, the magnitude of the output voltage also decreases.
However in the proposed circuit, the boost converter maintains the DC link voltage even if the generator
power drops; therefore, the output side can maintain its power.
Figure 18 shows the same analysis with the second condition, where the asymmetric format is applied.
As expected, the loss in the primary side remains unchanged; however, the loss in the secondary side is
reduced approximately 18% compared to that for the symmetric format in Fig. 17. When the battery is atfull power, the loss reduces approximately 25%. This comparison shows that the proposed converter canachieve better efficiency by applying the asymmetric method and the performance of the boost converter
is not limited by the change of carrier.
CONCLUSION
A new control method is proposed by utilizing the neutral point of a motor and connection to anindirect matrix converter for motor drive applications. Control over the inverter zero vector periods
allows an additional chopper leg to perform as a boost converter with connection to the neutral point of a
motor. Simulation and experimental results demonstrated good sinusoidal waveforms and confirmed thevalidity of the proposed method. From the loss analysis of the proposed circuit, an efficiency of 96% wasestimated. EMC behavior of the circuit will be subjected for further investigation.
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