High Efficiency Isolated DC/DC Converter
using Series Voltage Compensation
Jun-ichi Itoh, Satoshi Miyawaki, Nagaoka University of Technology, Japan
Kazuki Iwaya, TDK-Lambda Corporation, Japan

Abstract
A new circuit topology is proposed for a high efficiency isolated buck-boost DC/DC converter. The
proposed converter consists of a high efficiency resonance half-bridge converter and a series
converter. The proposed circuit regulates the voltage output by the series converter, which provides
only the differential voltage between the input voltage and output voltage. Therefore, the circuitachieves high efficiency when the input voltage is close to the output voltage, because only theresonance converter operates. The validity of the proposed circuit was confirmed by experiment andloss analysis, with a maximum efficiency of 95.8%.

1.

Introduction

Recently, the efficiency and size of power supplies for telecommunication applications and microprocessor boards have become very important issues. To achieve high efficiency and compact size, the power system is divided into
several converters, such as a point-of-load converter, bus converter and front end converter. In
this system, the power is distributed to the eachdevice by a DC voltage line [1-3].
The DC bus converter, which regulates theconstant DC voltage, is required at the input
point of each device, because the DC bus voltage fluctuates due to the bus impedance. The
DC bus converter is also used to obtain other DC
voltage levels from the DC bus voltage. High efficiency DC/DC converters are required for thebus converters.
Resonant type half-bridge converters, which
use the leakage inductance of transformers, are
one of the most effective circuit topologies to obtain high efficiency. However, it is difficult toregulate the output voltage range for an input
voltage fluctuation, because the switch timing isconstrained by the resonance period. Therefore,a resonant-type half-bridge converter is generally
connected to a voltage control converter, suchas a buck chopper [4-5]. As a result, the converter loss increases, because all the power
passes through two converters; the voltage control converter and a resonant-type converter.
This paper proposes a new topology for an
isolated DC/DC converter with series voltage
compensation. In the proposed circuit, the range

of input voltage fluctuation is compensated by an
auxiliary circuit that outputs only the differential
voltage between the input and output side [6].
One of the advantages of the proposed circuit isthat high efficiency can be achieved when the
input voltage is close to the output voltage, because the power of the auxiliary circuit becomesvery small in comparison to the input power.
Generally, the large fluctuations of the DC bus
voltage are not generated for long time periods,
and as a result, a decrease of the converter loss
can be achieved.
Firstly, the approach used to obtain high efficiency with the proposed series compensation
method is introduced in this paper. The circuit
configuration that achieves the proposed method
is described. Secondly, the operation mode isanalyzed using a simple equivalent circuit of the
proposed circuit. The stability of operation for the
proposal circuit is clarified. In addition, the indicator for optimum design is clarified. Finally, experimental results and a loss analysis are presented in order to demonstrate the advantagesof the proposed circuit.

2.

Proposed Converter

2.1.

Principle of series converter

Fig. 1 shows the configuration of a conventional DC/DC converter. The conventional circuit
consists of a resonant half-bridge converter anda buck chopper. The fluctuation of the input voltage is constantly controlled by the buck chopper.

The voltage is then isolated by the resonant halfbridge converter.
This system has a two times power conversion
from input to output. The total converter efficiency of the conventional circuit ηc is obtained
from (1), using both the resonant type converter
efficiency η1 and the buck chopper efficiency η2.
(1)ηc = η1η 2
All power passes though both converters, despite the relation between the input and output
voltage; therefore, the converter efficiency is decreased.
Fig. 2 shows the configuration of the proposed
converter. In the proposed circuit, the resonant
type half-bridge converter is used for the main
circuit that channels most of the power. Zero current switching (ZCS) can be achieved by using
the leakage inductance of the transformer andthe resonance capacitor in the DC part, in orderto achieve high efficiency.
The switching frequency of the main circuit isconstrained by the resonance frequency. Fromthe equivalent circuit of the transformer shown in
Fig. 2, the resonant inductance of the proposed
circuit L is obtained from (2), using the leakage
inductance lxx, mutual inductance Mxx and thenumber of wire turns Nxx.
M 1 (l12 + LR 2 )(2)
L = l11 +M 1 + (l12 + LT 2 )22

where LT 2 = ⎜⎛ N11 ⎞⎟ ⎛⎜ N 22 ⎟⎞ ⎜⎛ l21 + l22 M 2 ⎞⎟⎜N ⎟ ⎜N ⎟ ⎜⎝ 12 ⎠ ⎝ 21 ⎠ ⎝

1
2π LC

On the other hand, a full-bridge converter isused for the auxiliary circuit that controls the output voltage. The output voltage of the auxiliary
circuit is added to the output of the main circuitby the transformer. That is, the auxiliary circuit
compensates only the differential voltage between the input voltage and the output voltage
commands.
The proposed system consists of two power
converters, which are connected in series to the
output side. High efficiency is achieved in the
proposed circuit as follows. The power Pout is obtained by adding the auxiliary converter power P3
to the half-bridge converter power P1. The total
converter efficiency of the proposed circuit ηp is
obtained by (4), using the auxiliary circuit efficiency η3 and the power ratio k=P3/P1.
ηp =η1 + kη 3
1+ k

Vout

Fig. 1. Conventional circuit using a buck chopper and a resonant converter.

Fig. 2. Proposed circuit using series
compensation.
Therefore, the total efficiency ηp of the proposed circuit is higher than the conventional circuit when the relation of efficiency for each circuit
is given by (5).
ηp =

(3)

(4)

Resonant converterVinl22 + M 2 ⎟⎠The resonant frequency of the proposed circuit fo
is obtained using (3)fo =Buck chopperη1 + kη3
1+ k> η1η 2(5)

It should be noted that the proposed circuit
uses many switching devices. However, the current rating of the auxiliary circuit is much lower
than that of the main circuit.

2.2.

Control Strategy

In the proposed circuit, the main circuit is controlled under optimum conditions by the resonant
converter. In order to obtain maximum efficiency,
the resonant converter functions at a 50% duty
cycle, which is synchronous to the resonancefrequency fo. Therefore, the main circuit canachieve ZCS at any time.
Fig. 3 shows the operation mode and the
waveforms in secondary side of the transformer
when in boost mode. The switching timing of theauxiliary circuit is synchronous with the main circuit. The auxiliary circuit outputs three voltage
levels; +Vdc,–Vdc and zero voltage. When the input voltage is decreased (boost mode), the auxiliary circuit outputs a positive phase voltage tothe main circuit without preventing the resonance
operation of the main converter. Similarly, whenthe input voltage is increased (buck mode), theauxiliary circuit outputs a negative phase voltage.
The output voltage is controlled by changing
the pulse width D of the auxiliary circuit. Whenthe voltage drop from the leakage inductanceand the winding resistance is negligible, the output pulse width D is obtained by (6), using theswitching cycle T and the output voltage command Vout*.
*V ⎞T N ⎛ N(6)
D = ⋅ 21 ⎜ 12 − out ⎟2 N 22 ⎜⎝ 2 N 11
Vin ⎟⎠
Fig. 4 shows the control block diagram of the
proposed circuit. The duty cycle of the main circuit is set to 50%. In the auxiliary circuit, the output pulse width D is calculated by an automatic
voltage regulator (AVR). The phase shift gain isthen calculated according to D. The phase shift
is required to decrease the switching loss in theauxiliary circuit. According to the relation between the output and input voltages, the mode
selector determines the operation mode. Notethat the proposed converter has buck, boost or
reference voltage modes, so that the auxiliary
circuit does not operate when the input voltage isalmost the same as the output voltage.

Fig. 3. Correspondence of the switching patternand operation mode (boost operation).
Carrier
50 %

+

Phase
shift
gain cal.

AVR-

Duty 50 %+Phase
shifter

MUX

Phase
shifter

MUX

Sm1
Sm2
Sa1
Sa2
Sa3
Sa4

Mode
selecterFig. 4. Control block diagram.
2:1

C

R1

L1

ideal

±Vin

ideal

±Vout

iTm vTm

3.

Operation mode analysis

Sadd

R2

R1=r11+r12
L1=l11+l12
R2=r21+r22
L2=l21+l22=N11/N12=N21/N22

L2
ideal

3.1.

Simple equivalent circuit

Fig. 5 shows the equivalent circuit of the proposed circuit under boost operation. In AC signal
analysis, the switching operation of the main circuit and the rectifier part on the output side isexpressed as a square wave, which has the amplitude of the input and output voltage, and thefrequency of the switching frequency. The capacitors of the half-bridge converter can be represented as one resonance capacitor and idealtransformers with a turn ratio of 2:1. Moreover,the full-bridge converter in the auxiliary circuit isrepresented as two switches. When Sadd isturned on, the additional voltage mode is engaged. When Sshort is turned on, the zero voltage
mode is engaged. These switches function withmutual and complementary synchronization of
the switching frequency. Note that the excitationinductance of the transformer is disregarded, because it is much larger than the leakage inductance.

3.2.

SshortDerivation of the circuit equation

Fig. 6(a) and (b) show the simplified equivalentcircuit separated into the main circuit side andvTa

Fig. 5. Equivalent circuit of the proposed converter (boost operation).
iTm_m C

R1

L1

2

2

R2

L2
± Vout_m±

Vin(a) Main circuit only.
Sadd iTm_a C
±

Vin

R1

Sshort

L1

2

2

R2

L2
± Vout_a(b) Auxiliary circuit only.
Fig. 6. Simplified equivalent circuits for eachcircuit according to the superposition
principle.
auxiliary circuit side, respectively, according to
the superposition principle. The separation of theequivalent circuit is achieved by focusing on thetransformer input current iTm of the main circuit.
The equivalent circuit is obtained from the per-

spective of the main circuit, when the input of theauxiliary circuit is short-circuited. Similarly, theequivalent circuit is obtained from the perspective of the auxiliary circuit, when the input of themain circuit is short-circuited. iTm is obtained by(7),(7)
iTm = iTm _ m + iTm _ awhere iTm_m is the input current on the main circuit side shown in Fig. 6(a) and iTm_a is the input
current on the auxiliary circuit side shown in Fig.
6(b).
Consequently, the circuit equation of the entire
circuit can be obtained by deriving each equationfor each circuit equation in Fig. 6. The input currents iTm_m and iTm_a are obtained from (8) and (9).
⎡1 V⎤V
⎢ ⋅ in sin (ωt ) − α ⋅ out _ m sin (ωt )⎥ωL⎢ 2 ωL
⎥
1 ⎢
⎥
− t
⎧
⎫
1
⎥
sin (ωt )⎬iTm _ m = e τ ⎢+ im ( k ) ⎨cos(ωt ) −
⎢
⎥
2Q
⎩
⎭
⎢
⎥
⎛
1 ⎞
⎢
⎥
⎢− qm ( k )ω ⎜⎜1 + 4Q 2 ⎟⎟ sin (ωt )
⎥
⎝
⎠
⎣
⎦
(8)
⎤
⎡α V
Vout _ a⎢ ⋅ in sin (ωt ) − α ⋅sin (ωt )⎥ωL⎥⎢ β ωL
⎥
1 ⎢
− t
⎧
⎫
1
⎥
iTm _ a = e τ ⎢+ i a ( k ) ⎨cos(ωt ) −sin (ωt )⎬
⎥
⎢
2
Q
⎩
⎭
⎥
⎢
⎛
⎥
⎢
1 ⎞
⎟ sin (ωt )⎥⎢− q a ( k ) ω ⎜⎜1 +
2 ⎟
4Q ⎠
⎝
⎦
⎣
(9)

where τ = 2 L , ω =
R

ωL
1
⎛ 1 ⎞
,
−⎜ ⎟ , Q=
LC ⎝ τ 2 ⎠
R
2

2

⎛α ⎞
⎛α ⎞
R = R1 + ⎜⎜ ⎟⎟ R2 , L = L1 + ⎜⎜ ⎟⎟ L2 ,
⎝β ⎠
⎝β⎠

im(k) and ia(k) are the initial currents of the leakage inductions, and qm(k) and qa(k) are the initialcharges of resonance capacitor for the main andauxiliary sides, respectively.
Thus, the current vibrates at angular frequency
ω, and exhibits the phenomenon known as series resonance.

4.

Design method

4.1.

Basic principle and circuitspecifications

Fig. 7 shows the design flowchart for the proposed circuit. Optimization of the design isachieved by following the design flow. Firstly, thedesign specifications, which consist of the volume of the circuit, the minimum power of the stable operation area Pmin, the fluctuation range ofthe input voltage Vfluc, the input and output voltage Vin-Vout, and the rated power Pmax, are determined by application. The resonance frequency is determined by the switching frequency.
The resonance impedance is then designed
based on the specifications. Specifically, theminimum value of the resonance inductance L islimited by the minimum power of the stable operation area Pmin. After L is determined, C can beobtained by (3) according to the resonance frequency. Note that the resonance impedance isadjusted so that the maximum voltage of C is notlarger than the minimum input voltage.

4.2.

Determination of resonant inductanceIn order to obtain high efficiency, stable operation is required. The boundary condition forstable operation depends only on the operationof the auxiliary circuit (Fig. 6(b)). Instability iscaused by discontinuance of the resonance current in the light load area. In this state, the peakvalue of the transformer current increases significantly compared to that in continuous mode,
because the same output power is obtained asthat using a narrow current pass period. This is
the same problem encountered for the continuous or discontinuous current mode of a boost reactor in a chopper circuit. Therefore, the continuous mode has to be maintained until theminimum power Pmin is achieved. The lower limitof the light load in which continuous mode ismaintained is dependent on the resonance
impedance.
Fig. 8 shows the resonant current waveforms
for the separate auxiliary circuit with the boundary condition shown in Fig. 6(b). The boundary
condition indicates the edge state between thecontinuous and discontinuous current. At theboundary condition (i), the current value iTm_a until t1 is zero. Next, iTm_a in section (ii) can be obtained from (10) using a linear approximation according to the inductance L and its voltage. Notethat the capacitor voltage is equal to the output
voltage.
iTm _ a ( ii ) =

α Vin
⋅
⋅D
β L

(10)

iTm_a becomes the maximum at t2, and decreases in section (iii). The output voltage period(output pulse width D in the auxiliary circuit) islimited according to (11).

2 Do >

T
−D
2

(11)

VolumeI0 =⎫1 ⎧ 1 α Vin
D (D + D0 )⎬⎨ ⋅T 2 ⎩2 β L⎭

Vout _ a

(12)

(13)

αRL

Decision: L
Calculation: C

ExamplesCircuitSpecifications

48 V – 48 V
12 V (±25%)
75 W
200 W
250 kHzOn the other hand, the current mean value I0 is
obtained from the output power with (13), using
the load resistance RL and the output voltage
Vout_a.
I0 =Lower limit of L
Calculation

INPUT

where Do is the zero voltage period of the auxiliary circuit for a half cycle.
When the linear approximation is used, the current waveform can be approximated to a triangular shape. Therefore, the current mean value I0
for the half cycle is obtained by (12).

Decision: fo

Vc_maxCalculationNo
Vc_maxRating of Vc ?

Yes

Selected
component

Fig. 7. Design procedure flowchart.

Thus, the range of L for stable operation is obtained by (14), using (11) to (13).
L≥

α 2Vin RL D ⎛
D ⎞
⎜1 +
⎟
4 βVout _ a ⎜⎝ T 2 ⎟⎠(14)

Consequently, the minimum value of L is limitedby the output pulse width D and the load condition.

4.3.

Design example

The table in Fig. 7 shows the specifications forthe circuit.
The value of the minimum resonance inductance is obtained by (14) using the specifications
given in Fig. 7. When Pout = 75 W, approximately
2.5 µH is necessary for the resonance inductance. The inductance is inserted in series to theprimary side of the transformer in the main circuit,because the leakage inductance of the maintransformer is insufficient. Moreover, C is determined by (3) to be 0.19 µF.
The maximum value of the resonance current
Imax is then obtained from (15), using the maximum power Pmax.
I max = 2 ⋅Vinπ⋅ = 13A
V Pmax 2
2
out

(15)

Therefore, the maximum value of the resonance capacitor voltage Vc_max can be obtained
from (16).
I max
=45V
(16)
ωC
Vc_max is lower than the input voltage. Consequently, each element in the proposed circuit isselected.
Vc _ max =Fig. 8. Resonant current waveforms for the
separate auxiliary circuit with the boundary condition shown in Fig. 6(b).

5.

Experimental results

Fig. 9 presents the efficiency of the proposed
converter at constant load (load: 100 W, output
voltage: 48 V) when the input voltage has fluctuation of ±25%. To confirm the validity of theconcept, the proposed converter was tested. A
maximum efficiency of 95.8% was obtained, as
shown in Fig. 9, when the input voltage was very
close to the output voltage to maximize efficiency.
The reason for the lower efficiency of the buckmode compared to that of the boost mode is thatthe current is increased due to the circulation
current between the auxiliary and main circuits,
and the switching loss is increased due to an increase in the input voltage.
Fig. 10 shows the input current of the transformer and the terminal voltage of Sm2. In bothboost and buck modes, the half-bridge converter
can maintain zero current switching. A switchingfrequency of approximately 250 kHz was confirmed.
Fig. 11 shows the loss analysis based on thesimulated and experimental results. The simulated efficiency is in good agreement with that

6.

Conclusion

A series-type isolated DC/DC converter consisting of a resonant-type half-bridge converter
and an auxiliary converter was proposed. The
concept for series compensation is to obtain the
output voltage by adding or subtracting the auxiliary converter voltage to the half-bridge converter
voltage. The operational mode of the proposed
circuit was analyzed using a simple equivalentcircuit, and the optimal design approach wasclarified.
The experimental results demonstrated amaximum efficiency of 95.8% when the input
voltage was close to the output voltage. It was
confirmed that voltage control can be regulated
while maintaining high efficiency.
In future work, the circuit topology will be optimized to reduce the switch count in the auxiliary
circuit.

7.
[1]

[2]

[3]

[4]

[5]

[6]

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98
97

Efficiency [%]determined experimentally. Therefore, the converter loss can be discussed with respect to theloss analysis simulation. As a result, the switching loss in the auxiliary circuit increases in bothboost and buck modes, and the loss in the transformer and the rectification diode are predominant.

96
95
94
93

Output voltage : 48 V

92

Output power : 100 W

91

Switching frequency : 242 kHz

90

36

40

44
48
52
Input voltage [V]5660

Fig. 9. Efficiency of the proposed circuit (experimental result).
Sm2
Voltage

50 V/div
0

5 A/div

Trans.1
0Input current

1 ms/div(a) Boost mode (Vin = 40 V)
Sm2

50 V/div

Voltage0

5 A/div

Trans.1
0Input current

1 ms/div(b) Buck mode (Vin = 56 V)
Fig. 10. Waveforms of the input current of thetransformer and the terminal voltage Sm2.

8
7
6
5
4
3
2
1
0
1

2

3

Fig. 11. Loss analysis at 100 W load based onloss simulation.
series compensation", PESC 2008.IEEE, pp2109-2114, 2008.