AN50009

Power MOSFET applications in automotive BLDC and PMSM
drivesRev. 2.0 — 5 July 2022

application note

Document information
Information

Content

Keywords

BLDC, PMSM, motor drive, simulation

Abstract

This application note provides an overview of electrical machine use in vehicles. PMSM and BLDC
operating principles are detailed. A simple switch selection and switch loss estimation is provided
along with simulation results to verify the calculations. An accompanying interactive version of thisapplication note, including embedded simulations can be found on Nexperia.com

AN50009

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Power MOSFET applications in automotive BLDC and PMSM drives

1. Introduction
With today’s top of the range cars having more than 40 electrical machines, the global demand for
motor drives rapidly rises. This is especially true for Brushless DC (BLDC) and Permanent Magnet
Synchronous Motor (PMSM) drives. BLDC and PMSM are essential parts of Hybrid Electric Vehicle(HEV) and Electric Vehicle (EV) propulsion systems towards which priorities globally seem to be
directed. BLDC and PMSM drives are also employed where higher power and better regulationis needed in internal combustion vehicles. Some of the key automotive applications for electricalmachines and drives are depicted in Fig. 1. Motor applications can be implemented in any of thethree main categories shown:
1. Powertrain – Energy related – key aspect is Performance.
2. Chassis and Safety – Safety and Comfort – key aspect is Reliability.
3. Body control – Ease of use and Lighting – key aspect is Cost.
aaa-033950

Powertrain

Body Control• Window lifter (DC Motor)• Wiper (DC Motor / BLDC )• Seat control (DC Motor)

Chassis & Safety• Steering (BLDC / PMSM)• Sunroof (DC Motor / BLDC )• Power tailgate (DC Motor / BLDC )Fig. 1.

• Transmission (BLDC/ PMSM)• Cooling system (BLDC)• Start-Stop (BSG & ISG – BLDC / PMSM)• Hybrid drive (BLDC / PMSM (higher power
than Start-Stop))

Key automotive applications for electrical machines and drives

2. Electrical machines in automotive applications
Most of the electrical machines in vehicles are volume produced Brushed DC motors that do nothave complex speed and torque control requirements. They can be employed in applications suchas door locks, mirror folding, electrical seat adjustment and window motors.
The unidirectional and bidirectional Brushed DC motor drives are shown in Fig. 2. More information
about these drives can be found in application notes AN50004 and IAN50004.

Ma) unidirectional
Fig. 2.

AN50009

application noteMb) bidirectional

aaa-033951

MOSFET driven Brushed DC motor drive: a) unidirectional and b) bidirectional

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Power MOSFET applications in automotive BLDC and PMSM drives
Higher power and control complexity drives in today’s vehicles are mostly realised with BLDC
motors. Examples of such applications are water-pump, engine-cooling, anti-locking brake system,fuel-pump and electric steering. These motors, unlike the Brushed DC motors, do not need aphysical connection to the rotor. This enables greater robustness, less maintenance, higher powerand speed operations. Besides, as it can be seen from the drive configuration in Fig. 3 there
are more MOSFETs employed in the drive compared to the Brushed DC motor. This enables
current sharing across more devices, inherently increasing the power that can be delivered. As
it will be shown later on, the switches operate in sequence, so that one of the three switchingpairs does not operate at any instant, allowing the devices to cool. Also, the roles of the switching
MOSFET in one phase and the conducting MOSFET (not switching) in the other active phase canbe swapped. Both of the afore mentioned methods allow for better distribution of losses across the
six MOSFETs, in turn enabling higher power margin up to the maximum die temperature.

M

aaa-033952

Fig. 3.

Three-phase BLDC and PMSM drive

Applications that require even higher powers such as the drives involved in the propulsion of thevehicle are mostly realised with PMSM drives. Some of these applications are Electric Power
Steering, Starter/Alternator and Transmission pre-charge pumps. The noisier and higher torqueripple operation of the BLDC makes their employment in these applications undesirable. Whilethe higher efficiency, higher power and torque density makes investment in the more expensive
PMSMs justified for these applications.
Despite having the same drive structure as the BLDCs (shown in Fig. 3) the PMSMs havecompletely different modulation and control methods. In fact, even the machine structure is similar,
with the difference being in the shape of the produced Back Electro-Motive Force (EMF) having
trapezoidal shape with the BLDCs and near-sinusoidal with the PMSM machines.
In the area around a kilowatt of power, there is prospect for both the BLDC and PMSM applications
to be designed. Therefore, a 1 kW, 48 V, 3 phase system for the two machine types will be
investigated here.

2.1. PMSM drive theory
This section will explain in simple terms the main principles of PMSM drive. As the name suggests,
the PMSM has a permanent magnet (or an array of them) mounted on its rotor. The stator createsa Tesla’s rotating magnetic field. Thanks to the sinusoidal shape of the stator currents and a nearsinusoidal distribution of the stator winding, this field has a constant magnitude but its spatial angle
is changed so that it rotates with a uniform speed around the stator circumference, somewhat likethe arms of a clock. This magnetic field is the equivalent to having a magnet rotating along theinternal stator circumference. Naturally, the magnet of the rotor would tend to align with the stator
magnet, causing it to rotate along, at the same speed as the stator field, somewhat as it is shown in
Fig. 4.

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Power MOSFET applications in automotive BLDC and PMSM drives
aaa-033953

Fig. 4.

PMSM stator-rotor interaction

In order to achieve near-sinusoidal current in all machine phases, all three inverter legs are Pulse
Width Modulated (PWM) modulated with sinusoidal reference signals. Sensitive and expensive
encoders or resolvers are used to sense the exact rotor position in order to recreate the right phaseof the reference sinusoidal supply voltages.
When driven in generator mode, torque is applied to the rotor, forcing it to rotate. This in turnrotates the magnets on the rotor, creating a rotating magnetic field sensed by the stator winding.
Due to the sinusoidal spatial distribution of the stator magnets the rotating field induces sinusoidal
voltages in the stator windings. If a load is applied to the stator connectors a current will flow
through them. This can be seen in Simulation 1 of the interactive application note IAN50009 asshown in Fig. 5.

Fig. 5.

AN50009

application note

Simulation 1. PMSM in generator mode

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Power MOSFET applications in automotive BLDC and PMSM drives

2.2. BLDC drive theory
Similarly, to the PMSM the BLDC also has permanent magnets attached to its rotor. However,instead of a rotating field on the stator, the phases are subsequently abruptly energised to pull
the rotor magnets forward. This results in a somewhat jerky motion of the rotor, which is filtered tosome extent by the rotor and load inertia.
The principle is illustrated in Fig. 6.
I

a
S

c

T1

H2

1

Active switches

H1

b-

2
a+

3

4
b+

c-

5

a-

6
c+

1

b-

a+

S

Vdc
a

b

I

b

N

c

H3

Hall sensor
outputs

Vdc

H2

T2

a

b

b

H1

N
H3

H3
ea
ia

-I

-E

eb
Back-EMFs, 0Phase currents

S

I

H2

0

S

c

T1

0

0
E
I
0

a

I

H1

N

T6

2

Operating mode

N

1

ib
ec

c

0

ic
aaa-033363

Fig. 6.

BLDC motor operation principle
In the first instance, the third inverter leg is idle while the current flows from the top switch of thefirst inverter leg through phase ‘a’ and phase ‘b’ in reverse direction and finally to the negative dc
rail through the lower switch of the second leg. This corresponds to a south pole generated fromphase ‘a’ and a north pole generated from phase ‘b’. This driver state is assumed once ‘H1’ hallsensor starts sensing the north pole of the rotor.
Because of the construction of the stator and the rotor phase ‘a’ winding pulls the rotor north poleand pushes the rotor south pole away. This state is useful until just before the middle of the rotor
south pole reaches the stator phase ‘b’ winding. At this instant H3 hall sensor starts sensing therotor south pole and the drive configuration changes to the second state in Fig. 6 The currents
through the windings are now such that the north pole is generated with winding ‘c’ rather than ‘b’,
pushing the rotor north pole further along the stator inner circumference.
There are six such changes taking place in one revolution of the rotor, as shown on the rightsection of Fig. 6.

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Power MOSFET applications in automotive BLDC and PMSM drives

2.3. PMSM switch rating
When talking about a machine of certain power rating, it is assumed that we are talking about
the mechanical power that the machine is capable of delivering on its shaft. For determining theswitch rating of the drive, the input electrical power is needed. The difference between the input,
electrical and the output, mechanical power are the losses incurring within the machine and drive.
Throughout the machine, various losses appear due to magnetization, flow of electric current andmechanical movement of the rotor. These losses can be accounted for by the machine efficiency(η). Therefore, the electrical input power can be expressed as:

Pin = Pout

(1)

The efficiency of the machine depends on the machine type and design quality. For induction
machines of several hundred Watts it can go below 50%, while for high-power PMSMs it can reach95%.
The input power is then expressed as the product of the phase voltage and current root meansquare (rms) value:

Pin = 3V rms I rms Pf(2)

The number 3 signifies that the drive is three-phase, while the power factor (Pf) is a measure ofdisplacement in time of the voltage and current sinusoidal waveforms. The power factor, a measure
inversely indicating the proportion of energy needed for magnetisation of the machine, can also
take a broad range of values. PMSM have lower magnetisation requirement and therefore higher
power factor values compared to induction machines. They can be driven from unity down to 0.85
power factor.
An inverter is used to drive a PMSM with PWM output voltage. This is illustrated in Fig. 7 The
output voltage can take values of 0 V or Vdc. By applying longer pulses of Vdc the average value ofthe output voltage is increased. As the target (reference) output voltage should be of a sinusoidal
shape, with a negative and positive half-cycle, there is no other way of realising it than by adding
a dc bias of Vdc/2 to the reference voltage. This will not transfer to the current flow, it will just raise
the neutral point potential of the three-phase machine winding since the bias is applied to all threephases.

REFERENCE

Vdc

Load

INVERTEROUTPUT

4

6

8Time (ms)1012aaa-033364

Fig. 7.

Inverter leg with PWM output voltage

It is therefore clear that the maximal output voltage amplitude cannot be higher than Vdc/2.
However, a certain amount of third harmonic (and its multiples) can be added to the sinusoidal
reference. Once again, these harmonics are applied to all three phases, raising the voltage of all
three in the same manner and therefore no third harmonic current will flow.
The third harmonic can be added in such a way that it decreases the maximum of the reference,
allowing more headroom for the reference signal and increasing the modulation range (i.e. themaximum phase voltage applied) by an additional 15%. The same effect can be achieved by

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Power MOSFET applications in automotive BLDC and PMSM drives
calculating the average between the minimum and maximum references (out of the three available)
and adding it to all three reference signals. The described method can be studied within Simulation
2 of the interactive application note IAN50009 as shown in Fig. 8. If such reference signals areapplied, the performance will be equivalent to space vector modulation.

Fig. 8.

Simulation 2. Min-max injection modulation method

As the rms voltage is √2 times smaller than the maximum voltage in sinusoidal waveforms, it canbe obtained from the Vdc value as:

VV
Vrms = 1 dc 1.15 = 0.813 dc
2
2 2

(3)

Next, from Eq 2 the current rms value can be expressed. Finally, the maximal value of the current is√2 higher than the rms value:

I m = 2 I rms(4)

The chosen switch should have at least this current rating. In practise an overload factor of atleast 20% is added as well as a safety factor of another 100%. The highest voltage the switch
is expected to block is the supply voltage, in cases when the other switch in the same leg isconducting.

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Power MOSFET applications in automotive BLDC and PMSM drives

2.4. BLDC switch rating
BLDCs are mostly aimed at mid-power range applications where machine construction costs are
reduced, sacrificing some of the machine efficiency.
As it is shown in Fig. 6 and its description, one of the three inverter legs is dormant while two are
conducting. This means that the machine current along with a voltage up to Vdc/2 is applied to ardsingle phase of the drive 2/3 of the reference period. As there are three phases the input electric
power can be expressed as:

Pin = 3 2
3V dc
I = Vdc I
2(5)

Considering that the current is not as uniform as seen in Fig. 6 a waveform derating factor (wf) of
20% is used to obtain the maximum current the switch shall bear.

I m = 1.2 I(6)The maximum voltage that the switch should block is maintained at Vdc, as with the PMSM.
Table 1 presents a comparison of the required switch ratings for a 1 kW drive. As it can be seen,
the required current values of the switches are similar. The values for the voltage and current rms
for the BLDC can be obtained in a similar fashion as for the PMSM, with the ratio between therms and maximal values being instead of and without the 15% increase due to third harmonic
injection.
Table 1. Switch rating comparison for 1 kW drive
PSMN
BLDC

SourcePmeh

1 kW

Application
requirement

Vbat

48 V

Application
requirementPel

1.11 kW (η = 0.9)

1.17 kW (η = 0.85)

Eq 1

Vrms

19.51 V19.59 V

Eq 3

Irms

21 A (pf = 0.9)

24.5 AEq 2 and Eq 5

Im29.8 A

29.4 A (wf = 1.2)

Eq 4 and Eq 6

3. PMSM and BLDC losses calculation
The expected current and voltage that the switches will endure is calculated in the previous
section. However, as it is usually the case, the choice of MOSFETs is more constrained by theamount of losses they are designed to endure. The amount of losses is influenced by design
requirements (conducted current, switching speed and frequency) or cooling arrangements of thedevices.
In this section an explanation of how to calculate these losses will be provided. It needs to be
noted that usually at motor control applications the switching frequency is set to be as low as
possible as there is no incentive to have it higher. Most applications settle at 20-30 kHz, in order
to avoid audible frequencies. It is beneficial to have the switching at lower frequencies so that its
harmonics fade out on the frequency axis by the frequency where regulatory requirements start. By
having the switching frequency low, the switching losses are not of great relevance in motor control
applications.

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Power MOSFET applications in automotive BLDC and PMSM drives

3.1. PMSM conduction losses
The PMSM motor current is sinusoidal, with a certain switching frequency ripple superimposed. Let
the period of the sinusoidal control signal be noted with Tc, while the switching period with Ts.
The current in positive direction will flow through the top MOSFET in the positive control half-cycle(Tc /2) during the duty cycle δ in each switching interval Ts. In the negative control half-cycle (Tc /2)
for a period (1-δ) Ts negative current will pass through the same switch. Since the top and bottom
switches swap roles between control half-cycles (Tc /2) it can be concluded that during a wholecontrol cycle (Tc) one control half-cycle worth of current passes through the switch. This discussion
is illustrated in Simulation 3 of the interactive application note IAN50009, see Fig. 9.

Fig. 9.

Simulation 3. PMSM drive switch current conduction
The current direction is not relevant because the square of the current is sought. The overall
conduction losses can be calculated as:

Pcond = RDSon I 2rms

(7)

Where the rms current value is of a half cycle, averaged over the whole cycle period:2
I rms = 12∫0 i ( )d(8)This is valid if the reverse voltage drop of the device is below the internal diode conduction
threshold and dead-time is not accounted for. For a sinusoidal current with an amplitude Im, the[1]
conduction losses come to :2

AN50009

application noteI
Pcond = RDSon m
4(9)

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Power MOSFET applications in automotive BLDC and PMSM drives

3.2. BLDC conduction losses
In BLDC drives the current flows through two inverter legs. In order to regulate the current flowing
through the machine windings, at least one of the inverter legs needs to be modulated in order tomaintain a desired current magnitude. When only one inverter leg commutates and the other isconstantly conducting, the switching method is unipolar, when two legs commutate the switching
method is bipolar, as shown in Fig. 10. The unipolar switching shown in Fig. 10 distributes theswitching losses between the two FETs of an inverter leg. In both cases the upper and lower switch
are being turned on in an alternating manner, to avoid diode loses in the synchronous FET.
Bipolar switching
Operating mode

1

2

3

4

Unipolar switching
5

6

1

2

3

4

5

6

Phase a
A+

A-

A+

A+

A-

A+

Phase b
B-

B+

B-

B-

B+

B-

Phase c
C+

C-

C+

C+

C-

C+

aaa-033365

Fig. 10. Bipolar and unipolar switching of devices
For the conduction losses, the switching mode does not seem to have much influence. The
required rms current through the individual switches is the same in both cases. In the bipolarswitching mode, looking at the A+, switching at a certain duty cycle δ, a current of value I passes
through it for a time δTs (Ts being the switching period) in the first half cycle. In the second halfcycle A- is switched with duty cycle δ, while A+ is then switched on for (1 - δ)Ts. Therefore, during
the hole control cycle a single switch conducts the current equivalent of half a control cycle. Similar
is the case with the unipolar switch mode, with the difference that half of the switching cycle thesame switch conducts inherently.
Applying Eq 8 for a constant current I passing through the switch for one third of the switching
period, the resulting conduction losses come to:

Pcond = RDSon I
3

2

(10)

This conclusion applies when dead-times are very short and the current through the BLDC
windings is well controlled to be close to DC with small ripples.
The unipolar method can be simplified, having only the top switches driven with PWM, while thebottom switch is turned on inversely for freewheeling. The respective bottom switches, belonging
to the other conducting phase, are held constantly on during the time the current needs to flow
through them. In this case uneven distribution of power losses is achieved in the top and bottom
switches. The top switch is on for δTs. The bottom switch conducts when it is freewheeling for(1 - δ) Ts and for an additional third of the control period when it is held completely on. In this casethe integral from Eq 8 results in:

Pcond_up = RDSon I
3

2

2

Pcond_dn = RDSon I ( 1+ 1 - )3AN50009

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Power MOSFET applications in automotive BLDC and PMSM drives
In all the switching methods, the diode of the MOSFETs will need to demagnetise the motor
phase that stops conducting. The energy accumulated in the phase inductance will therefore bedissipated in each MOSFET diode in each control cycle:

E = 12 L I

2

(13)

Where L is the machine phase inductance.

3.3. Switching losses
The switching energy losses can be estimated as:

E on = Vdc Ion
E off = Vdc I ont sw(on)2(14)

t sw(off )2

Where tsw can be approximated by:t sw(on) =sf QGDR
Vgd - Vpl(Ion) Gt sw(off) =sf QGDR
Vpl(Ion) G(15)

Where tsw is the switching time, Ion is the on-state current, Vgd is the gate drive voltage, and RG is
the gate resistance. QGD – gate to drain charge and Vpl(Ion) – gate plateau voltage at current Ion,
can be read from the data sheet: QGD from tables and Vpl(Ion) from the transfer characteristic graph(Fig. 11).
240
ID
(A)
160

80
Tj = 175 °C
Tj = 25 ° C

Ion
0

0

2

4

V pl(Ion)

6

VGS (V)

8

Fig. 11. Reading Ion and Vpl(Ion) from the transfer characteristic
In itself, QGD accounts for the voltage transient during the Miller plateau. The current transientthat occurs before (turn on) or after (turn off) the Miller plateau is determined by part of QGS. It is
accounted for by an approximated increase of the voltage transient by an additional 20 to 30%,
represented by the scaling factor sf (sf takes value from 1.2 to 1.3). This percentage is dependent
on technology used. The approximation is sufficient here as the switching losses are not expected
to dominate because of the low switching frequency.

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Power MOSFET applications in automotive BLDC and PMSM drives
The on-state current is the average current in the case of the switching losses. For PMSM theaverage is calculated for half of the control cycle:

I avg = 1

∫ 0 i ( )d=Im

∫0 sin( )d

=

Im

(16)

This value can therefore be substituted in Eq 14 instead of Ion to obtain the switching energies. It isconsidered that the current ripple is negligible compared to the base value.

Psw =

f sw
( E on + E off )2(17)

In case of the BLDC the switching occurs around I, which needs to be placed in Eq 14:

I avg = I(18)

he obtained energy levels then need to be multiplied once again with the number of switchingrd
occurring in one half of the control period. Due to the device switching only 1/3 of the half-cycle(Fig. 10 – Unipolar switching) this is expressed as 1/3 × 0.5 × fsw/fcc.

Psw =

f sw
( E on + E off )6(19)

The switching at the negative half cycle is not accounted for in both cases as it is governed bydiode switching and it can be omitted. The reason for this is that by the nature of the diode thevoltage across it needs to decrease to nearly zero before it can take on any current, achieving
nearly zero voltage switching. The diode also cannot start increasing the voltage across its
connectors until the current has stopped flowing through it.

4. Switch selection
As noted, MOSFETs are usually chosen according to the amount of loss they are intended todissipate.
Based on low switching frequency – 20 kHz is chosen – the conduction losses are expected to
be dominant. At this point switching losses are estimated as 50% of conduction losses in PMSMs
and 20% in BLDCs. With an allowance of 1.5% losses in the six switches the required RDSon canbe calculated from Eq 9 and Eq 10 as shown in Eq 20 and Eq 21, respectively. The maximum
expected current amplitude can be read from Table 1:

0.015 Pin
RDSon = 42 Pcond , Pcond = 1
6
1.5
Im

(20)

0.015 Pin
RDSon = 32 Pcond , Pcond = 1
6
1.2
Im

(21)

This results in a RDSon of 8.3 mΩ for PMSM and 8.5 mΩ for the BLDC. To allow for some headroom
the BUK7Y7R8-80E is chosen. This is an automotive, 7.8 mΩ, 80 V, trench 6 technology MOSFET.
The drain current of this MOSFET is given as 100 A, roughly 3 times higher than the maximum
current needed for the application. This is usual and expected, as the ID rating of the MOSFETs
is measured with its mounting base temperature held at 25 °C. Table 2 gives an overview of the
expected switch performance for a gate drive voltage of 10 V and a gate resistance of 22 Ω.

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Power MOSFET applications in automotive BLDC and PMSM drives
Table 2. Expected switch performance
PSMN
SwitchAN50009

application note

BLDC

BUK7Y7R8-80E,
RDSon (typ) = 5.8 mΩ; VDS =80 V; ID = 100 A, QGD = 17 nC

SourceData sheet

Pcond

1.29 W

1.16 WEq 9 and Eq10

Ion

9.94 A

24.5 AEq 16 and Eq18

Vpl(Ion)

4.7 V

5V

Data sheet / Fig 8.

tsw_on, tsw_off

88 ns, 99 ns

19.59 V

Eq 15

Eon + Eoff

21 µJ, 23 µJ

55 µJ, 55 µJ

Eq 14

Psw

476 mW

375 mW

Eq 17 and Eq 19

Ploss

1.77 W

1.54 W

Pcond + Psw

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Power MOSFET applications in automotive BLDC and PMSM drives

5. PMSM and BLDC simulations
This section includes the drive loss simulation results from interactive application note IAN50009,
for the PMSM (Simulation 4) and BLDC (Simulation 5 and Simulation 6).
In all simulations switches for one inverter leg have been monitored for energy losses. The
electrical motor has been represented by its back EMF, phase resistance and leakage inductance.
In the case of the PMSM only one inverter leg is modelled. For the BLDC simulation the switches inthe two other inverter leg are replaced by ideal switches to shorten simulation time.
The total energy losses, the MOSFET conduction, turn on and turn off losses are then plotted andtheir end values displayed. These values should be divided by the simulation length (20 ms) toobtain the results from Table 2.
The results from simulation and calculations are compared in Table 3. Simulation 6 is not includedin the comparison as the top and bottom switches have a different losses distribution between
conduction and switching losses.
Table 3. Simulated switch performance
PSMN

BLDC

Simulation results

Theoreticalresults

Simulation results

Theoreticalresults

SwitchBUK7Y7R8-80E: RDSon (typ) = 5.8 mΩ; VDS = 80 V; ID = 100 A, QGD = 17 nC

Vcond

26.6 mJ / 20 ms = 1.33 W

1.29 W

23.8 mJ / 20 ms = 1.19 W

1.16 W

Psw(3.5 mJ + 7.8 mJ / 20 ms = 565 mW

476 mW(2.7 mJ + 4.3 mJ) / 20 ms = 350mW

375 mW

Ploss

1.89 W

1.77 W

1.54 W

1.535 W

Fig. 12. Simulation 4. PMSM simulation with loss estimation

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Power MOSFET applications in automotive BLDC and PMSM drives

Fig. 13. Simulation 5. BLDC simulation with loss estimation; unipolar modulation; all switches PWM modulated

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Power MOSFET applications in automotive BLDC and PMSM drives

Fig. 14. Simulation 6. BLDC simulation with loss estimation; unipolar modulation; top switches PWM modulated,bottom switches kept on

6. Conclusion
In this application note an overview of electrical machine use in vehicles has been given. A moredetailed look is then provided into PMSM and BLDC operating principles. A simple switch selection
and switch loss estimation is provided along with simulation results to verify the calculations.

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Power MOSFET applications in automotive BLDC and PMSM drives

7. References
1. J. W. Kolar, H. Ertl and F. C. Zach, "Influence of the modulation method on the conduction andswitching losses of a PWM converter system," in IEEE Transactions on Industry Applications,vol. 27, no. 6, pp. 1063-1075, Nov.-Dec. 1991, doi: 10.1109/28.108456.
2. Nexperia interactive application note IAN50009.

8. Revision history
Table 4. Revision history
Revision
Date
numberDescription

2.0

2022-07-05

Equation 17 and Equation 19 corrected.

1.0

2021-10-19

Initial version.

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Power MOSFET applications in automotive BLDC and PMSM drives

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Power MOSFET applications in automotive BLDC and PMSM drives

List of Tables
Table 1. Switch rating comparison for 1 kW drive................ 8
Table 2. Expected switch performance.............................. 13
Table 3. Simulated switch performance............................. 14
Table 4. Revision history....................................................17

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Power MOSFET applications in automotive BLDC and PMSM drives

List of Figures
Fig. 1. Key automotive applications for electricalmachines and drives............................................................ 2
Fig. 2. MOSFET driven Brushed DC motor drive: a)unidirectional and b) bidirectional........................................ 2
Fig. 3. Three-phase BLDC and PMSM drive....................... 3
Fig. 4. PMSM stator-rotor interaction................................... 4
Fig. 5. Simulation 1. PMSM in generator mode................... 4
Fig. 6. BLDC motor operation principle................................5
Fig. 7. Inverter leg with PWM output voltage....................... 6
Fig. 8. Simulation 2. Min-max injection modulation method. 7
Fig. 9. Simulation 3. PMSM drive switch current
conduction............................................................................ 9
Fig. 10. Bipolar and unipolar switching of devices............. 10
Fig. 11. Reading Ion and Vpl(Ion) from the transfer
characteristic...................................................................... 11
Fig. 12. Simulation 4. PMSM simulation with loss
estimation........................................................................... 14
Fig. 13. Simulation 5. BLDC simulation with loss
estimation; unipolar modulation; all switches PWM
modulated...........................................................................15
Fig. 14. Simulation 6. BLDC simulation with loss
estimation; unipolar modulation; top switches PWM
modulated, bottom switches kept on................................. 16

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Power MOSFET applications in automotive BLDC and PMSM drives

Contents
1. Introduction................................................................... 2
2. Electrical machines in automotive applications........ 2
2.1. PMSM drive theory......................................................3
2.2. BLDC drive theory....................................................... 5
2.3. PMSM switch rating.....................................................6
2.4. BLDC switch rating......................................................8
3. PMSM and BLDC losses calculation...........................8
3.1. PMSM conduction losses............................................ 93.2. BLDC conduction losses........................................... 10
3.3. Switching losses........................................................ 11
4. Switch selection......................................................... 12
5. PMSM and BLDC simulations................................... 14
6. Conclusion.................................................................. 16
7. References.................................................................. 17
8. Revision history..........................................................17
9. Legal information........................................................18© Nexperia B.V. 2022. All rights reserved

For more information, please visit: http://www.nexperia.com
For sales office addresses, please send an email to: salesaddresses@nexperia.com
Date of release: 5 July 2022

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