Methods for Reducing Leakage Electric Field of a
Wireless Power Transfer System for Electric Vehicles
Masaki Jo, Yukiya Sato, Yasuyoshi Kaneko, Shigeru Abe
Graduate School of Science and Engineering
Saitama University
Saitama, Japans13mm217@mail.saitama-u.ac.jp
Abstract—A wireless power transfer system for electricvehicles has to deal with the issue of high leakage levels ofelectric field. In this paper, we propose two methods for
reducing the leakage electric field. First is the use of ferrite to
reduce the emitted leakage electric field to the environment.
Second is the changing of the pulse width of the inverter toreduce the harmonics of leakage electric field. We present theexperimental results of the reduction effects of leakage electricfield. We also compared the leakage electric field using seriesparallel and series-series resonant capacitor topologies.

I.

INTRODUCTION

The development and commercialization of electricvehicles (EVs) and plug-in hybrid EVs (PHVs) are actively
being pursued because of environmental concerns andincreasing oil prices. Current PHVs and EVs requireconnection to a power supply for battery charging using
electric cables. A wireless power transfer system would havenumerous advantages: convenient, safe, and maintenancefree. Therefore, wireless power transfer systems are being
investigated worldwide [1], [2].
A wireless power transfer system for EVs must have alarge air gap and good tolerance to misalignment in thelateral direction. In addition, it must be compact andlightweight. The H-shaped core transformer has beendeveloped to satisfy these requirements [3], [4]. However, it
has the issue of leakage electromagnetic field emission fromthe transformer to the environment. The leakage magnetic
and electric fields should be regulated because they may
negatively affect the human body and electronic devices. For
the effect on the human body, the guideline is defined in theInternational Commission on Non-Ionizing Radiation
Protection (ICNIRP) 2010 [5], and the leakage of the 3-kW
H-shaped core transformer should not exceed the reference
level at the outside of the vehicle. On the other hand, withregard to the effect on electronic devices, the electric field
intensity emitted from an H-shaped core transformer exceeds
the reference value set for induction cookers by the Radio
Law in Japan. Therefore, the H-shaped core requires some
types of countermeasures. However, very few reports

978-1-4799-5776-7/14/$31.00 ©2014 IEEE

Figure 1.

Methods for reducing the leakage electric field.

concerning the issue of reducing leakage electric field areavailable.
In this paper, we propose two methods for reducing theleakage electric field intensity (Fig. 1). Here, the leakage
electric field is defined as the equivalent electric field, whichis calculated as E = Z × H, where E is the electric field, H is
the measured magnetic field, and Z is the impedance in freespace (120 π). Therefore, reducing the magnetic field isnecessary to reduce the equivalent electric field. The first
method uses ferrite to reduce the propagation of the leakage
electric field. The second method employs pulse width
control of the inverter to reduce the harmonic current at the
primary side and the leakage electric field. We compared theprimary magnetomotive force with the secondary one andshowed that reducing the harmonic current at the primaryside is more effective. We confirmed the effectiveness ofboth methods by an experiment that measured the leakage
electric field intensity. We also compared the leakage electricfield using series-parallel (SP) and series-series (SS) resonantcapacitor topologies.

1762

Figure 2.
I'IN

Wireless power transfer system.
r'1

-jx'S

jx'1

jx2

r2

I2

Figure 5.

TABLE I. TRANSFORMER SPECIFICATIONS
Type
H-shaped core

jx'0
-jxP

V'IN
r'0

Figure 3.

Outline of an H-shaped transformer.

ID

RL

V2

Primary

Weight[kg]Detailed equivalent circuit.

Winding

6.70

Secondary

6.43

Primary

3p×20T

Secondary

15p×4T

TABLE II. TRANSFORMER PARAMETERSType
H-shaped core(a) Simplified equivalent circuit
Figure 4.

II.

(b) Ideal transformer(a) Simplified equivalent circuit.
(b) Ideal transformer.

PRIMARY AND SECONDARY CURRENTS OF A
TRANSFORMER

The equivalent leakage electric field depends on themagnetomotive force NI, which is the product of the numberof turns N and the winding current I in a wireless powertransfer system. Therefore, examining the magnitude andphase angle of primary current IIN and secondary current I2 isimportant. Fig. 2 shows a schematic diagram of thecontactless power transfer system using series and parallel
resonant capacitors. A full-bridge inverter is used as a highfrequency power supply.
Fig. 3 shows the detailed equivalent circuit, which
consists of a T-shaped equivalent circuit with resonantcapacitors CS and CP and load resistance RL. The primaryvalues are converted into equivalent secondary values using
the turn ratio a = N1/N2. Because the winding resistances andferrite-core loss are considerably lower than the leakage andmutual reactances at the resonant frequency, the winding
resistances (r'1 and r'2) and ferrite-core loss r'0 can be ignored(Fig. 4 (a)).

50

r1 [mΩ]

136

r2 [mΩ]

5.25

l0 [μH]28.4

l1 [μH]117

l2 [μH]

4.87

CS [μF]0.0722CP [μF]

1.70

k

0.192

b

0.189

RLmax [Ω]10.3

ηmax [%]

97.0

The input voltage V'IN and the input current I'IN can be
expressed as
′ =
V IN

V IN
x 0′
I
′ = aI IN = L , b =.
= b V 2 , I INab
x 0′ + x 2

The relationship between IIN and I2 can be expressed as⎛ x ′ − x1′Z′ ⎞⎟ I I′N = acI IN+j
I 2 = ⎜⎜1 − S
⎟
′
x
x
0
0′ ⎠
⎝

⎛
x′ − x′Z′ ⎞⎜⎜ c = 1 − S 1 + j ⎟⎟ .
′
x0
x0′ ⎠
⎝

1
= x P = ω 0 L2 = x 0′ + x 2 ,
ω0 CP

(1)

(2)

Equation (2) shows that the equivalent circuit of atransformer with these capacitors is the same as that of an
ideal transformer with turn ratio b at the resonant frequency(Fig. 4 (b)).

The secondary parallel capacitor CP and the primaryseries capacitor CS can be expressed as

1x′ x= xs′ = 0 2 + x1′ .
ω0CS′
x0′ + x2

f0 [kHz](3)

where Z' is {x'0/(x'0+x2)}2. From (3), we can determine
whether IIN or I2 causes more leakage electric field.

1763

III.

LEAKAGE ELECTROMAGNETIC FIELD IN AN H-SHAPEDCORE TRANSFORMER

1000
500
flux density [μT]

A. Transformer Specifications
The leakage magnetic and electric fields should be
regulated because they may negatively affect the humanbody and electronic devices. ICNIRP provides the reference
levels for human body, and the Radio Law in Japan provides
the reference levels for electronic devices. In this section, we
compare the electromagnetic field with the specified
reference levels and investigate the necessity of decreasing
the electromagnetic field. The specifications of thetransformer used in our experiment are shown in Fig. 5 andTable I and II.

C. Leakage Electric Field
The leakage electric field may negatively affectelectronic devices. Because no clear standard of leakage
electric field is currently set for wireless power transfer
system for EVs, we examined the leakage electric field using
the criteria for induction cookers as a reference, which isdefined by Article 46-7 of the Enforcement Regulations of
the Radio Law in Japan (JP-46-IH). The reference level isdetermined for a distance of 30 m, and because ourmeasurement was at 3 m, we converted the reference level tothe 3-m value (106 dB μV/m, except for the operating
frequency). The measurement of the electric field intensity
was performed under the following conditions: 1.5 kW, 50
kHz, mechanical gap length of 150 mm with nomisalignment, and distance of 3 m in a fully anechoic room.
A 60-cm-diameter loop antenna (ETS-Lindgren Model 6512)
was used in the measurement.
Fig. 7 and Table III show the leakage electric field
intensity of the H-shaped core transformer. Compared withthe criteria for induction cookers, the electric field intensitiesof the H-shaped core transformer are approximately 26, 14,
and 3 dB over the criteria at 50, 150, and 250 kHz,

100
50 27
10
5
1
0.5
0.1

300
1000
distance from center [mm]Figure 6.

B. Leakage Magnetic Field
The leakage magnetic flux may negatively affect thehuman body. The reference level of the leakage flux density(27 μT at 50 kHz) is set by ICNIRP 2010. Fig. 6 shows theleakage flux density of the H-shaped core transformer with amechanical gap length of 150 mm without misalignment andwith a 3-kW output power. Owing to the directionality of the
H-shaped core transformer, the measurement is performed in
the forward direction x and lateral direction y. The decreaseof flux density in the x direction can be expressed as B ∝ r-2.7,
whereas that in the y direction can be expressed as B ∝ r-2.5(B is the flux density, and r is distance from the center of thetransformer).
The leakage flux densities of the H-shaped coretransformer in the x and y directions are below 27 μT at
distances of 650 and 500 mm from the center of thetransformer, respectively. Thus, the H-shaped coretransformer can be considered safe from outside a vehicle if
the vehicle width is 1700 mm and the transformer is attachedto the vehicle center.

H-shaped(x)H-shaped(y)E field intensity [dB μV/m]

160

Flux density (POUT = 3 kW).

H-shaped(x)H-shaped(y)

140

3000

H-shaped(x)H-shaped(y)JP-46-IH(fundamental)
JP-46-IH(spurious)

120
100
80
60
10

Figure 7.

100Frequency [kHz]Electric field intensity (POUT = 1.5 kW).

respectively. Thus, reducing the electric field intensity in the
H-shaped core transformer is important.
IV.

METHOD FOR REDUCING THE LEAKAGE ELECTRIC
FIELD

Two methods are available to reduce the leakage electricfield. One is to control the propagation path of the leakage
electric field by shielding the electromagnetic waves emitted
from the transformer. The other is to control the generation
source of the leakage electric field by reducing the harmonic
current flowing in the transformer.
The former method investigates the reduction effect of
the leakage electric field using a ferrite shield placed at theback of an aluminum sheet. The latter method investigates
the harmonic reduction effect of the leakage electric fieldusing pulse width control inverter.
A. Propagation Path Control
1) Reduction effect of ferrite shield
To reduce the leakage electric field externally radiatedfrom the transformer, we propose a method that uses a ferrite
with a high magnetic permeability at the back of analuminum sheet. Fig. 8 shows the sizes of the ferrite and thealuminum sheet. Considering the weight of the onboard-side
device, the ferrite shield is used only at the primary side. The
experimental conditions are the same as those described in
the previous section.

1764

TABLE III. ELECTRIC FIELD INTENSITY USING FERRITE SHIELD

Before countermeasures

Al sheet[mm]

50 kHz150 kHz250 kHz350 kHz1.5

450 × 450

146.2

119.6

108.9

103.2

1.5

450 × 450

143.3

113.3

100.6

94.6

1.5

600 × 600

142.0

112.4

100.9

95.5

Plate

1.5

600 × 600

143.4

114.8

103.4

97.5

Frame

1.5

600 × 600

144.2

116.2

104.5

99.1

-

-

120

106

106

106

H-shaped
Plate + Frame

Propagation pathcontrol

Electric field intensity[dB μV/m]

POUT
[kW]

Type

JP-46-IH (3m)

155

E field intensity [dB μV/m]

H-shaped
145

Plate+Frame

JP-46-IH(fundamental)

135
125

Plate
Frame

JP-46-IH(spurious)

115
105
95
85
50

150

250

350

Frequency [kHz]Figure 8.

160

E field intensity [dB μV/m]

Figure 10. Electric field intensity using ferrite shield.

Dimensions of the transformer and ferrite shield.

H-shaped

140 H-shaped(with ferrite)
120

aluminum sheet to increase the distance between the ferrite
shield and the magnetic poles. By changing the aluminum
sheet size from 450 mm × 450 mm to 600 mm × 600 mm
and the ferrite shield size from 490 mm × 510 mm to 650
mm × 660 mm, the coupling factor k increased from 0.135 to
0.165.

H-shaped
H-shaped(with ferrite)
JP-46-IH(spurious)

100

Fig. 10 and Table III show the measurement result of theleakage electric field intensity. The electric field intensity
was reduced by approximately 7.2, 8.1 and 7.7 dB at 150,
250 and 350 kHz, respectively. The reduction effect wasalmost the same as that of the 490 mm × 510 mm ferrite
shield, and the transformer efficiency improved from 90.1%
to 92.7%.

80
JP-46-IH(fundamental)

60

10Figure 9.

100Frequency [kHz]Electric field intensity using ferrite shield.

Fig. 9 and Table III show the measurement result of theleakage electric field intensity. The leakage electric field
intensity with the ferrite shield was lower than that without
it. The electric field intensity was reduced by approximately
2.9, 6.3, 8.4, and 8.6 dB at 50, 150, 250, and 350 kHz,
respectively. However, the coupling factor k decreased from
0.192 to 0.135, and the transformer efficiency decreased
from 93.1% to 90.1%.
2) Improvement in efficiency
The main flux was reduced because the magnetic flux
generated by the primary side was not coupled to thesecondary side but to the ferrite shield since the distance
between the ferrite shield and the magnetic poles of the
primary transformer was close. Therefore, the transformer
efficiency was reduced when the ferrite shield was used. To
solve this problem, we considered increasing the size of the3) Ferrite amount reduction
The use of ferrite for shielding has become increasingly
costly. To reduce ferrite usage, we studied the leakage
electric field in cases where only a ferrite plate and only aferrite frame are individually used. Fig. 10 and Table III
show the measurement result of the leakage electric field
intensity. Compared with that without any countermeasure,the leakage electric field intensity with only the ferrite plate
was reduced by approximately 4.8, 5.6, and 5.7 dB at 150,
250, and 350 kHz, respectively. That with only the ferrite
frame was respectively reduced by approximately 3.4, 4.5,
and 4.2 dB. The reduction effects in both cases were lower
than those when the plate and frame were used together.
By performing magnetic field analysis of the ferrite
shield using JMAG software, we found that the magnetic
flux density of the ferrite at the back of the aluminum plate is1765

Because the reactances of capacitors CS and CP in termsof the harmonics are small, the relationship between N1IIN
and N2I2 is approximately given by
N 2 I 2 ≈ bN1 I IN .

The c values in (3) at 50, 150, 250, and 350 kHz were
calculated as 1.1 0.22, 0.20, and 0.19, respectively.
Substituting the values of c in (4) yielded approximately thesame primary and secondary magnetomotive forces at an
operating frequency of 50 kHz. However, the primarymagnetomotive force was approximately five times larger
than the secondary one at 150 kHz or higher. Hence,
primary-side control of the generation source is more
effective than secondary-side control. Thus, we used a pulse
width control inverter for the primary-side harmonic current
reduction.

(a) Plate + frame(b) Reduction in the ferrite
Figure 11. Reduction in the ferrite amount.

10
Flux density [μT]52) Primary-side generation control
When a square-wave inverter is used as a high-frequency
power supply, the input voltage has a harmonic component

1
0.5

0.1
1000

H-shaped
Plate+frame
Plate
Frame
Reduction in the ferrite

δ

E

3000
distance from center [mm]θ0Flux density (reduction in the ferrite).

relatively low. Therefore, we performed the analysis for thecase where the ferrite amount in the x–y plane was reduced
by approximately 62%, as shown in Fig. 11(b). Fig. 12shows the simulated result of the magnetic flux density 3 mfrom the center of the transformer. The magnetic flux density
for the case of the ferrite with a 62% reduction was lower
than those when only the ferrite plate and only the ferrite
flame models were individually used. The reduction effect ofthe magnetic flux density for the case of the ferrite with a
62% reduction was similar to that when the frame and plate
were combined. From these results, we can possibly reduce
the ferrite amount by approximately 62% despite a slightdecrease in the effect.

−E

Figure 13.

H-shaped
Pulse width control
SPL

155
145
135
125
115
105
95

50

150

B. Generation Source Control

250

350

Frequency [kHz]The relationship between IIN and I2 is expressed by (3).
Because a = N1/N2, the relationship between N1IIN and N2I2 isgiven by
(4)

1766

(a)

Primary winding

165

H-shaped
Pulse width control
SPL

155
145

I2[dBμA]

1) Comparison of magnetomotive forces
The equivalent leakage electric field intensity is
proportional to the magnetomotive force NI. The transformer
is divided into the ground-side primary coil and onboard-side
secondary coil, and we do not know which side has more
effect on the leakage electric field. This section first presents
which side, whether the primary or the secondary side can beeffectively controlled for the generation source by
calculating and comparing the primary and secondary
magnetomotive forces N1IIN and N2I2.

N2 I 2 = cN1I IN .

Pulse width control inverter.

165

IIN[dBμA]

Figure 12.

(5)

135
125
115
105
95

50

150
250
Frequency [kHz](b) Secondary winding
Figure 14. Harmonic currents.

350

TABLE IV. ELECTRIC FIELD INTENSITY WITH GENERATION SOURCE CONTROL
Type
Beforecountermeasures

H-shaped

Pulse width
control
SPL
Propagation path andFerrite shield
generation sourceand pulse
controlwidth control
JP-46-IH (3m)Generation source
control

E field intensity[dB μV/m]

POUT
[kW]

Al sheet[mm]

50 kHz150 kHz250 kHz-350 kHz1.0
1.5

450 × 450
450 × 450

146.0
146.2

117.1
119.6

107.0
108.9

101.5
103.2

1.0

450 × 450

145.9

107.6

107.3

98.1

1.5

450 × 450

145.1

119.7

108.4

102.0

1.0

600 × 600

141.6

88.5

100.4

90.6

-

-

120

106

106

106

155

H-shaped

H-shaped
Pulse width control

E field intensity [dB μV/m]

E field intensity [dB μV/m]

160
140 Pulse width control

JP-46-IH(spurious)
H-shaped
Pulse
width control

120
100
80

JP-46-IH(fundamental)

60

H-shaped145

JP-46-IH(fundamental)

135
125

Pulse width control
Plate + Frame and Pulse
width control

JP-46-IH(spurious)

115
105
95
85

10

100
Frequency [kHz]

50

150

250

350

Frequency [kHz]Figure 16. Electric field intensity.

Figure 15. Electric field intensity (Pulse width control).

whose magnitude is inversely proportional to the order. For
this reason, the primary current includes the harmonics. We
adopted a pulse width control inverter. Fig. 13 shows thepulse width control wave-form. Pulse width δ was set to avalue that can reduce the leakage electric field at the third
harmonic of 150 kHz.
Fig. 14 shows the simulated result of harmonic currents
IIN and I2. IIN was reduced by approximately 15 dB at 150
kHz. Figs. 15 and 16 and Table IV show the measurement
results of the leakage electric field intensity. The
experimental conditions were the same as those described in
Section 3 except that the output power was 1.0 kW. The
electric field intensity under the pulse width control became
approximately −9.6, +0.3, and −3.3 dB at 150, 250, and 350
kHz, respectively, compared with that without the control.
3) Secondary-side generation control
To verify that the effect of the secondary-side generation
control is small, the experiment was performed using an SPL
topology that inserted reactor LS before the full-wave
rectifier [4].
When a full-wave rectifier is used, the current that flows
through the rectifier has a pulsed waveform that flows onlyaround the peak of the rectifier input voltage. Therefore, I2
includes the harmonics. The use of the SPL topology can
suppress the harmonic current that flows to the rectifier
circuit and can reduce the harmonic component of I2.

Figure 17.

L inserted at the secondary side.

Fig. 17 shows the schematic diagram of the SPL
topology. LS is determined to have the same value as L2, asshown in (6).
ω 0 L S = ω 0 L 2 = x 0′ + x 2(6)

Fig. 14 shows the simulated result of harmonic currents
IIN and I2. I2 was reduced by approximately 11 and 14 dB at
250 and 350 kHz, respectively.
Table IV shows the measurement results of the leakage
electric field intensity of the SPL topology when the output
power is 1.5 kW. The electric field intensity using the SPL
topology became approximately +0.1, −0.5, and −1.2 dB at
150, 250, and 350 kHz, respectively. The reduction effect of
the SPL topology was low, although the harmoniccomponent of I2 was reduced.
4) Pulse width control using ferrite shield
By combining the two methods, i.e., when the ferrite
shield and pulse width control of the inverter were used, theelectric field intensity was reduced by 4.6, 28.6, 6.6, and 10.9
dB at 50, 150, 250, and 350 kHz, respectively (Fig. 16 andTable IV).

1767

160

E field intensity [dB μV/m]

E field intensity [dB μV/m]

160

SP topology
SS topology

140

SP topology
SS topology

140
120

120

100

100
80
60
10

50 100frequency [kHz]

80
60
10

500

50 100frequency [kHz]500(b) y direction(a) x direction
Figure 19. Electric field intensity of the SP and SS topologies.
160

148.0 149.7

SP topology
SS topology

140

E Field intensity [dB μV/m]

E Field intensity [dB μV/m]

160120.7 122.3

120110.1 108.2

104.6

100

97.6

80
60
40

147.4 147.9

SP topology
SS topology

140118.5 120.7

120107.9 107.3

100

101.8

98.1

80
60
40

fundamental(84.2kHz)

3rd
5th(254.5kHz)(423.8kHz)
frequency

7th(594.1kHz)

fundamental(84.2kHz)(a) x direction

3rd
5th(254.5kHz)(423.8kHz)
frequency

7th(594.1kHz)(b) y direction

Figure 20. Comparison of electric field intensity of fundamental and harmonic components.
TABLE V. TRANSFORMER PARAMETERSType
SP topology
SS topology
Winding

4p×14T
15p×4Tf0 [kHz]

Figure 18.

Schematic diagram of the SS topology.

From these results, we confirmed that the proposed
methods are effective in reducing the leakage electric field
intensity.
V.

LEAKAGE ELECTRIC FIELD INTENSITY OF THE SP AND
SS TOPOLOGIESThe SP and SS topologies are the most common
capacitor compensation topologies. In this section, we
compare the SP and SS topologies when the operating
frequency is 85 kHz and the output power is 3 kW. Fig. 18
shows the schematic diagram of the SS topology. The
primary series capacitor CS1 and secondary series capacitor
CS2 values that resonate with each self-inductance value aredetermined as follows:
1
′
ω0CS1

′ = x0′ + x1′ ,
= xs1

1

ω0CS2

= xs2 = x0′ + x2 .

(7)

4p×14T85

r1 [mΩ]

85.8

81.7

r2 [mΩ]

7.55

77.4

l0 [μH]

13.7

14.3

l1 [μH]

58.4

57.3

l2 [μH]

4.62

56.5

CS [μF]0.05040.0491CP [μF]

0.611

0.0497

k

0.192

0.199

RLmax [Ω]

15.5

7.3

ηmax [%]

97.4

97.9

Table V lists transformer parameters of the SP and SS
topologies. Figs. 19 and 20 show the leakage electric field
intensity in the x and y directions. The leakage electric field
intensity is proportional to the sum of the primary and thesecondary magnetomotive forces. From the experimental
results, the magnetomotive force of the SP topology was 368
A, and that of the SS topology was 399 A. The

1768

magnetomotive forces were approximately the same, and theleakage electric field intensity was at the same level.
VI.

REFERENCES[1]

CONCLUSION

The H-shaped core transformer for wireless powertransfer system has to deal with the issue of high levels ofleakage electric field intensity. We have proposed two
methods for reducing the leakage electric field intensity
using ferrite as a propagation path control and pulse width
control of the inverter as a generation source control. Whenthe ferrite was used, the leakage electric field intensity was
reduced by approximately 2.9, 6.3, 8.4, and 8.6 dB at 50, 150,
250, and 350 kHz, respectively. When the pulse width
control of the inverter was used, the leakage electric field
intensity at 150 kHz was reduced to approximately 9.6 dB. A
further countermeasure against the leakage electric field at an
operating frequency of 50 kHz is necessary. We also
confirmed that the leakage electric field intensity of the SP
and SS topologies are at the same leakage level.

[2]

[3]

[4]

[5]

[6]

1769

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