Switched flyback PFC converter
for wide AC input voltage range
Yuki Kawai1, Naoto Kikuchi1, Hiroki Watanabe1, Keisuke Kusaka1, and Jun-ichi Itoh1
1

Nagaoka University of Technology, Japan

Abstract— This paper proposes a switched flyback PFC
converter for a universal input range applications. Conventional
controllers have large number of components to realize the wide
range voltage input, which reduces the efficiency. The proposed
converter supports worldwide power supplies by two flyback
converter and two additional switches. The proposed converter isbased on two flyback converters and switches the connections ofthem in series or parallel according to the input voltage.
Moreover, the proposed circuit is driven by the Triangular
Current Mode (TCM) with variable frequency (VF) to achievezero-voltage switching (ZVS). However, the VF causes the input
current distortion because the input current is not proportional tothe input voltage by changing the duty ratio. Therefore, this paper
proposes the compensation method of the input current distortion
using the TCM. The fundamental operation is demonstrated by
the simulation and experiment. As the simulation result, it was
confirmed that the input current distortion was improved by
99.2%. In the experimental result, the proposed circuit drives thepower factor of 99.9 % by 1.0 p.u. in DCM operation.
Index Terms—Flyback converter, Power
Correction, Wide range, Zero Voltage Switching.

FactorI. INTRODUCTION
Isolated AC/DC converters have been widely used as
the switching power supply in many applications to obtainthe DC voltage from single-phase or three-phase grid [15]. Power Factor Correction (PFC) techniques have been
widely considered to compensate for the input current
distortion to satisfy the IEC standard [6-8]. The two-stage
configuration using the boost-type PFC converter is thecommon solution to obtain the sinusoidal input current [911]. However, additional magnetic components andswitching devices are required for the PFC function,
deteriorating the conversion efficiency [11,12]. In addition,
the inductor in the boost circuit increases the volume of theconverter. On the other hand, the single-stage PFC
converter is also considered due to attractive advantages
such as the simple configuration and the low number ofcomponents [13,14].
High power density and high-efficiency power
conversion are also challenges for the isolated AC/DC
converter with the universal input range [15-18]. The
converters with the switching capability of theseries/parallel connection have been proposed to extend
the high-efficiency range with the various AC input
voltage [19-23]. The parallel operation reduces current
RMS and the conduction loss. On the other hand, the series
operation reduces the voltage stress of switching devices.
However, these converters require additional switching
devices to change the series/parallel connections.

In order to solve the problem, the PFC converter basedon the flyback converter has been proposed [11,24-26].
The flyback converter has advantages such as simple
circuit configuration, low cost, galvanic isolation between
input and output, and buck-boost operation capability. The
flyback converter can improves the power factor of the
input current without additional stages by operating with
the Discontinuous Current Mode (DCM), which keeps theduty ratio and the switching frequency constant, to control
the input current [27].
Zero Voltage Switching (ZVS) [28-30] using the
Triangular Current Mode (TCM) and using the quasiresonant method improve the efficiency of the flyback
converter. The quasi-resonant method generates theresonant current to discharge the energy of the parasitic
capacitance of the switching device during the dead-time
period. On the other hand, in the TCM operation, themagnetizing current is controlled to the negative todischarge the parasitic capacitor. However, these ZVS
methods are necessary to vary the switching frequency. As
a result, the input current quality deteriorates because thesemethods do not meet the operating conditions for the PFC.
This paper proposes a switched flyback converter for
the wide-range AC input voltage with the series/parallel
connection. The proposed circuit consists of two flyback
converters and a changeover circuit. The contribution ofthis paper is the realization of the high-power density andhigh efficiency in the wide range input voltage. Moreover,the control method is proposed to improve the input
current quantity using the TCM operation by canceling thetime variation of the duty ratio. As a result, the simulation
results reveal that the proposed control method reduces the
input current harmonics by 99.2 % compared to theconventional method. The validity of the proposed method
is demonstrated by the simulation and experiment.
This paper is organized as follows. First, in section II,
the circuit configuration of the proposed circuit isintroduced and the operation is clarified. Next, in section
III, the design of the controller and circuit parameters aredescribed. In section IV, the validity of the proposed
method is confirmed by the simulation. In section V, theproposed method is verified by the 240-W experimental
prototype. Finally, the paper concludes in section VII.
II. CIRCUIT CONFIGURATION OF SWITCHED FLYBACK PFCCONVERTER

Figure 1 shows the circuit configuration of the proposed
switched flyback converter. The proposed circuit consistsof a synchronous rectifier and two active clamp flyback
converters, and two additional switching devices for series

and parallel operation. The active clamp flyback converter
has the capability of PFC, galvanic isolation, and DC
voltage control for Vdc. The active clamp circuit whichapplies ZVS absorbs the surge voltage due to the leakage
inductance of the high-frequency transformer. The
proposed circuit adds two additional switching devices of
S6 and S7 to realize the series and parallel operation.
The series and parallel operations are used dependingon the input voltage variations. The series operation isapplied under the high input AC voltage condition todivide the input voltage into two flyback converters. On
the other hand, the parallel operation is used to share thelarge input current when low input AC voltage conditions.
III. OPERATION MODE
A. Series Operation
Figure 2 shows the operation mode of the proposed
converter with a series connection. The proposed converter
has two switching modes for the primary side of the
flyback converter. The series operation composes the Input
Series Output Parallel (ISOP) configuration. The universal
input requires a high voltage and current rating for the
power converter because the input specification is widely
changed depending on the applications. Therefore, it isdifficult to optimize the circuit design to realize the higher
power density and efficiency when one converter is used.
The series operation based on ISOP is used for highinput AC voltage conditions such as 300Vrms to 600Vrms.
The primary side of two flyback converters is connected inseries to divide the input AC voltage. The secondary sideof the flyback converter is connected in parallel to reduce
the large current RMS in each converter because thesecondary voltage is low as 24 V.
The series operation has two switching modes as shownin Fig. 2. Mode 1 starts when S7 is turned on to charge themagnetic energy to the magnetizing inductance of two
high-frequency transformers. Note that the magnetic
energy is also charged to the leakage inductance Lleak1 andLleak2.
Mode 2 starts after mode 1 through the dead time. In thismode, the magnetic energy of the magnetizing inductanceis transferred to the load when S9 and S10 are turned on. In
this mode, S11 and S12 are turned on to absorb the huge
surge voltage in S7 due to the leakage inductance. The
active clamp circuit achieves ZVS to reduce the switchingloss.
B. Parallel Operation
Figure 3 shows the operation mode of the proposed
converter with a parallel connection. This operation also
has two modes same as the series operation. The parallel
operation composes the Input Parallel Output Parallel(IPOP) configuration to divide the input current. This
operation is used for low input AC voltage conditions as
85 Vrms. Mode 3 starts when S5 and S8 are turned on to
charge the magnetic energy to the magnetizing inductance.
In this mode, each magnetizing inductance is connected inparallel. Finally, mode 4 starts after mode 3 through thedead time. This mode is for transferring the magnetic
energy to the load same as mode 2.

i1 ileak
Lleak
S1

S2 C1

Lf

vin

S11

S9

im
Lm

S5
S6

Cf
S3

S4

C2

S7

AdditionalswitchSynchronousrectifier

Lleak

S10

Lm

Vdc

S12

S8

Two Active clamp
flyback converter

Fig. 1. Circuit configuration of proposed converter.
Lleak
C1

Lm

S11

S9 S

10

S5
S6
S7

Lleak
C2

Lm

S12

S8
Switching state: ON: S6,S7
OFF:S5,S8,S9,S10 ,S11 ,S12(a) Mode1
Lleak
C1

Lm

S11

S9 S

10

S5
S6
S7

Lleak
C2

Lm

S12

S8

Switching state: ON: S6,S9,S10 ,S11 ,S12
OFF:S5,S7,S8(b) Mode2
Fig. 2. Switching state of series operation.
Lleak
C1

Lm

S11

S9 S

10

S5
S6
S7

Lleak
C2

Lm

S12

S8
Switching state: ON: S5,S8
OFF:S6,S7,S9,S10 ,S11 ,S12(a) Mode3
Lleak
C1
S11

Lm

S9 S

10

S5
S6
S7

Lleak
C2

Lm

S12

S8
Switching state: ON: S6,S9,S10 ,S11 ,S12
OFF:S5,S7,S8(b) Mode4
Fig. 3. Switching state of parallel operation.

IV. CONTROL METHOD

NVdc....................................................... (4)M (t ) =Vin sin( t )where Vin is the amplitude of the input voltage, Vdc is theoutput voltage, Ton is the actual On-state period, and M isthe voltage ratio between primary and secondary side
voltage, respectively. In the TCM control, the switchingfrequency varies depending on the grid voltage. In the caseof the constant frequency modulation with the
Discontinuous Current Mode (DCM), the sinusoidal input
current is obtained when the constant duty ratio is applied.
On the other hand, the input current in TCM is distorted
due to the non-linear characteristics. Therefore, thelinearization term (t) is added to compensate for the input
current distortion. (t) is multiplied to the On-statecommand Ton* to eliminate the non-linear term in (3). The
linearization term is given by
1 + M (t )............................................................. (5) (t ) =M (t )On the other hand, the turn off time Toff(t) for thesecondary side switch of S9 and S10 is given by
Vin sin(t )Lleak
Toff (t ) =(Ton* − Td ) ............... (6)
Lm + Lleak N (Vdc + V f )where Td is the dead time, Vf is the forward voltage of the
body diode in secondary switching devices. Besides, theperiod-to-period variation of Ibot is very small that it isassumed to be negligible.

Ton

Magnetizing current im [A]Tde la y

Toff

0-Ibot*
Gate signals(Series: S7 Parallel: S5,S8)

Gate signals(Series, Parallel: S9,S10 ,S11 ,S12 )Gate signals(Series: S6 Parallel: S6,S7)
S6: Always ON (Series ope.)
S6,S7: Always OFF (Parallel ope.)
Series, Parallel:
S9, S10,S11,S12
ZVS

Series: S7 Parallel: S5,S8

ZVS
Drain-source voltage [V]Fig.4. Operation principle of transformer current anddrain-source voltage.
DSP

FPGA

Ton *

×

Figure 4 shows the magnetizing current waveform,
switching state, and drain-source voltage of the flyback
converter. The control method is implemented based on
the Triangular Current Mode (TCM) to achieve ZVS for
the flyback converter. The magnetizing current increases
during mode 1 and mode 3 in each series and parallel
operation. During the dead time, all switching devices areturned off. In this period, the parasitic capacitances of S9,
S10, S11, S12 are discharged by the magnetizing current.
And then the magnetizing current decreases during mode
2 and mode 4 until the zero current. in TCM, the switchingstate of mode 2 and mode 4 continue during the delay
period Tdelay to make the negative current as shown in Fig.4.
The parasitic capacitance of S5, S6, S8, is discharged by thenegative current during dead time. As the result, ZVS isachieved.
Figure 5 shows the control block diagram of the
proposed control method. The TCM control isimplemented based on the Boundary Current Mode
(BCM) control. Firstly, the magnetizing current increases
during the On-state period Ton*. And then the magnetizing
current decreases until zero current. In the case of the BCM
control, the switching state is changed to Ton* when thezero current is detected. On the other hand, the TCM
control continues the same switching state to generate thenegative current. This switching state remains during thedelay period Tdelay*.
The average of the primary current ipri_avg (t) using the
TCM is given by
V sin(t ) M (t )M (t )(3)
ipri_avg (t ) = in
Ton − I bot
2 ( Lm + Lleak ) 1 + M (t )1 + M (t )vin_det ABS
Vdc _de t
Detection

Cal. of
 (t)Eq.(5)

Ton
Gate 8
Pulse
Gen. S5~12



Tde la y*
im_de tDetection

ZCD
ZCD: Zero current detection

Fig. 5. Control block diagram of proposed circuit.
Table. 1. Simulation parameters.
SymbolOutput power
Output DC voltage
Input voltage
Turn ratio
Magnetizing inductance
Grid frequency

P
Vdc
vin
N:1
Lm
f

Value
240 W
24 V
200,600 Vrms
4:1
0.22 mH
50 Hz

V. SIMULATION RESULT
Table 1 shows the simulation parameters. The input
voltage is 600Vrms in the series operation, the input voltage
is 200Vrms in the parallel operation, and the rated power is240W. Note that the active clamp circuit and the input
filter are not considered in this simulation.
Figure 6 shows the operation waveform compared withthe conventional and proposed method in parallel
operation. In the conventional method, the THD of the
input current is 20.5%. On the other hand, in the proposed
method, the THD is 0.17%.

Figure 7 shows the operation waveform compared withthe conventional and proposed method in the series
operation. In the conventional method, the THD of the
input current is 24.3%. On the other hand, in the proposed
method, the THD is 3.24%.
Figure 8 shows THD characteristics in the proposed
circuit. As shown in figure 8, the input current THD of theseries and parallel operation are drastically improved inwide output power condition. Especially, the input current
THD of the parallel operation was improved to 20.1%
from 1.57% when the output power is 0.1 p.u.. According
to this result, it was confirmed that the input current THD
was improved in wide output power condition by the
proposed control.

VI. EXPERIMENTAL RESULT
Table 2 shows the experimental parameters. In thisexperiment, a series connection and a parallel connection
are used at the input voltage 300Vrms, 150 Vrms respectively.
Note that the diode rectifier and the capacitor are usedinstead of the synchronous rectifier.
Figure 9 shows the experimental results in the proposed
circuit using DCM operation. Fig.(a) shows theexperimental result of a series connection. Fig.8(b) showsthe experimental result of a parallel connection. As theexperimental result, the input power factor is 0.99 in aseries connection and a parallel connection. In addition,
the operation waveform of the primary current is generated
during the zero-current period.
Input voltage : vin [V]Input voltage : vin [V]
200

500
0

0

-500

-200
0.5

0.02 s/div

0.02 s/div

Input current : iin [V]

0.2

0.0

Input current : iin [V]

0.0

-0.5

THD 20.5 %-0.2(a) Without proposed control

THD 20.3 %(a) Without proposed control

Input voltage : vin [V]Input voltage : vin [V]

200

500

0

0
-500

-200

0.02 s/div

0.02 s/div
Input current : iin [V]Input current : iin [V]
0.2

0.5
0.0

0.0

-0.5
-0.2
THD 0.17 %(b) With proposed control
Fig.6 Simulation result of parallel operation.

THD 3.24 %(b) With proposed control
Fig.7 Simulation result of series operation.

VII. CONCLUSION
This paper proposed the switched flyback PFC
converter to realize the high-power density and highpower factor for a wide AC input voltage range. The
proposed converter reduces component counts by sharingof switching devices of each converter. In addition, theproposed converter divides the voltage and current
condition by the series and parallel connection.
Furthermore, the ZVS method to ensure compatibility with
PFC capability was proposed. The validity of the proposed
method was demonstrated by simulation and experiment.
As the simulation result, the input AC current THD was
improved by 99.2 % when the proposed control wasapplied. As the experimental result, the validity of the
proposed circuit is confirmed using the DCM operation.
Thereby, the THD of the input current was 6.06 % in aseries connection at 1.0p.u.. And, the THD of the input
current was 1.54 % in a parallel connection at 1.0p.u..
In the future work, the validity of the proposed control
with TCM will be considered in the experiment.

25
25

1.0p.u = 240W.

Input current THD [%]Figure 10 shows the THD characteristics in the
proposed circuit. The flyback converter compensates forthe harmonic component of the input current. However, inlight load, THD of the input current exceeded 10% withboth controls. Improving the THD is future work.

Red line: Series operation
Black line: Parallel operation

18.3% reduction

15
15

Proposed method (Vin=600V)
Conventional method (Vin=600V)
Proposed method (Vin=200V)
Conventional method (Vin=200V)

18.6% reduction
10
10

55

00

00

0.2
0.2

0.4
0.6
0.4
0.6
Output power [p.u.]

0.8
0.8

1.0
1

Fig. 8. THD characteristics compared with theconventional and proposed method.
Table. 2. Experimental parameters
SymbolOutput power
Output DC voltage
Input voltage
Turn ratio
Magnetizing inductance
Leakage inductance
Magnetizing inductance
Leakage inductance
Clamp capacitor
Input capacitor
Input inductor

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P
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Lm1
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8.948 mF
5 mF
1 mF
2.2 mH

Input voltage vin 250[V/div]

4.0[ms/div]00Input current iin 2.5[A/div]

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Series (vin=300V)
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