Wireless Power Transfer Systemwith the Auxiliary Resonant Commutated Pole
Converter for Reducing Radiated Emission
Rintaro Kusui
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japankusui@stn.nagaokaut.ac.jp

Keisuke Kusaka
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
kusaka@vos.nagaokaut.ac.jp

Hiroki Watanabe
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japanwatanabe@vos.nagaokaut.ac.jp

Jun-ichi Itoh
Dept. of Electrical, Electronics, andInformation Engineering
Nagaoka University of Technology
Nagaoka, Japan
itoh@vos.nagaokaut.ac.jp
Abstract— Wireless power transfer systems improve theconvenience for EV users. However, the WPT systems generate
the radiated emission caused by current harmonics thatinterfere with wireless communications. To solve this problem,
this paper proposes a WPT system with an auxiliary resonant
commutated pole (ARCP) converter with selective harmonic
elimination (SHE) on the primary and secondary sides. The
ARCP converter suppresses the increasing switching losses on
main devices, and the SHE-PWM reduces coil current
harmonics components that cause radiated emissions.
Furthermore, the proposed system controls a constant boost
current to minimize losses in the auxiliary circuit. In this paper,
the switching phases are numerically analyzed in order toeliminate the low-order harmonics voltage of the converter
output voltage. Then, the operation of the proposed system isdemonstrated by simulation and experiment. As a result, it isconfirmed that the harmonics below the 13th order are reducedby more than 40 dB in the simulation. In addition, theexperimental results with the prototype of the three-pulse SHEPWM mention that all switching achieved zero voltage
switching.
Keywords— Wireless power transfer, Auxiliary resonant
commutated pole converter, Selective harmonics elimination,
Current harmonics, Radiated emission

I. INTRODUCTION
Wireless power transfer (WPT) systems for easy and safe
charging of electric vehicles (EVs) are actively studied [1-3].
The WPT system transfers power through a large air gap,
resulting in weak magnetic coupling. As a result, radiatedemissions are generated from the transmission coil. Radiatedemissions interfere with other electronic devices or wirelesscommunications and can adversely affect health. The
International Special Committee on Radio Interference(CISPR: Comité International Spécial des Perturbations
Radioélectriques) and the International Commission on NonIonizing Radiation Protection (ICNIRP) establish reference
levels for radiated emissions from transmission coils [4-5]. In
particular, the standard for the WPT system in CISPR 11 ClassA Group 2 is planned to be about 30 dB stricter for low-orderharmonic components [6]. For this reason, the WPT systems
require a reduction of the coil current harmonics that cause theradiated emissions.

XXX-X-XXXX-XXXX-X/XX/$XX.00 ©20XX IEEE

Most conventional WPT systems have a full-bridge
inverter on the primary side and a diode rectifier on thesecondary side as power supplies [7]. These power converters
output square wave voltages that include large low-orderharmonics. The square wave voltage applied to thetransmission coil and compensation circuit causes currents
containing harmonics to flow according to the frequency
characteristics of the resonator. The transmission coils
generate large leakage flux depending on the currents on coils
since the magnetic coupling between coils is very low. As aresult, the conventional WPT systems radiate the leakage
magnetic fields with large harmonic components [2].
The WPT systems with asynchronous PWM inverters andmulti-level converters, such as flying capacitor topology, havebeen proposed in order to reduce the low-order harmonics ofthe radiated emissions of the WPT systems[8-9]. The PWM
and multi-level converters reduce the output voltage
harmonics of the primary and secondary converters. As aresult, the harmonics of radiated emissions from thetransmission coils are reduced. However, almost all switching
operation becomes hard-switching. Thus, the losses and surge
voltages cause an increase in higher-order harmonic radiation.
In addition, the asynchronous PWM induces low-frequency
beat components in the output current and power.
Synchronous PWM avoids the output current and output
power beat components with selective harmonic elimination(SHE) [10]. However, the switching losses by hard switching
and radiation noise by surges are unavoidable.
Meanwhile, an auxiliary resonant commutated pole
(ARCP) converter has been proposed to reduce switchinglosses and radiation noise in power converters[11-12]. The
ARCP converter achieves zero-voltage switching (ZVS) in all
switching of the main devices by applying a resonant current
to the main switches just before commutation. As a result, the
ARCP converter suppresses the increase in switching losses
and the surge voltages in the main switches. Thus, the ARCP
converter can reduce radiated emissions. However, theauthors do not know that the ACRP converter has been applied
to high-frequency inverters, such as the WPT system, in orderto reduce radiated emissions.
This paper proposes a WPT system that has an ARCP
converter with SHE on the primary and secondary sides. The

new contribution of this paper is that the proposed system
reduces switching losses by ARCP and low harmonics
components of the inverter output voltage by SHE. This paper
demonstrates the reduction of the coil current harmonics
components by the simulation and the commutation operation
under the ZVS in the proposed system by the experiment.
II. PROPOSED WPT SYSTEM

B. ARCP Full-bridge converter
Figure 2 shows the output voltage and output current
waveforms and the switching timing of the ARCP full-bridge
converter during the commutation period. The resonant
inductor Lr is charged by turning on the auxiliary switch of the
ARCP just before the commutation of the main switches. The
charged current discharges the charges stored in the parasitic
capacitor Cr of the main devices. In this paper, the resonant
inductor current ir is minimized in order to minimize the lossesof the auxiliary circuit. The inductor charging time is adjusted
depending on the load current iL to keep the constant boost
current Ib. The inductor charging time tb1 in Fig. 2(a) and (b)is expressed as (1), and the charging time tb2 in (c) and (d) is
expressed as (2).

Lr iL + I b
VDC

Cr Sv11

()

Cr

C1

ir1 L1

L2

Sa12
Cr Sv12Cr Sv21

Cr

Sa21

Lr1

v1

Su21

C2

Sa11
VDC1

Su1 2

A. Circuit configuration of the proposed WPT system
Figure 1 shows the circuit configuration of the proposedsystem. The proposed WPT system has ARCP converters on
the primary and secondary sides. The resonant capacitors of
the WPT system are connected in series between thetransmission coil and the ARCP converter for both theprimary and secondary sides, which is called series (S-S)
compensation. The ARCP converters for this system achieve
ZVS by flowing a resonant current between phases in thecommutation period by applying unipolar modulation. Due tothis, the ARCP converter needs only one additional inductor
and two auxiliary switches. In addition, the proposed system
must have a short transient time to output a PWM-controlled
voltage with 85-kHz fundamental frequency for the WPT. For
this purpose, the ARCP of the proposed system uses only theparasitic capacitance of the main switch without the additional
resonance capacitor connected in parallel with the main switch
in the conventional ARCP.

tb1 =

Su1 1

VDC2

Lr2

ir2

v2

Sa22
Cr

i1

i2

Su22

Cr Sv22

Cr

Fig. 1. Configuration of proposed wireless power transfer system.
ir
iL

iL+I b
iL
ir

iL Ib

vo ut

vo ut

Sa2

Sa1

S1

S1
S2

S2
td _ma x

tb

td _mintd _min(a) S1 turn-on at iL>0tb(b) S2 turn-on at iL<0
iL
0i

Ib+iL
0
iL

td_max

tb

tb

ir

r

i L Ib

vo ut

vout

Sa2

Sa1

S1

S1

S2

S2
td _ma x

tb

td _ma x

tb

td _mintd _mintbtb(c) S1 turn-on at iL<0(d) S2 turn-on at iL<0
Fig. 2. Operation waveform of ARCP converter.

Cr

ir

Cr
Lr

vout
iL

Cr

Cr

Fig. 3. Equivalent circuit when the ARCP is in resonance condition.
VDC

VDC

0

b1

b2 b3

b4

bp

p/2

Fig. 4. p-pulse PWM waveform for Fourier series expansion (1/4 cycle).

tb 2 =

Lr I b − iL
VDC

()

where Lr is the inductance of the resonant inductor of the
ARCP and VDC is the DC link voltage. After the resonant
inductor is fully charged, turning off the main device initiates
resonance between the resonant inductor and the parasitic
capacitor of the main device. The negative current flows on
the other main switch by resonance. ZVS is achieved byturning on the main device during negative current flow.
Figure 3 shows the equivalent circuit of the ARCP
converter at resonance. From Fig. 3, the resonance current ir
and the inverter output voltage vout at resonance in Fig. 2(a)are expressed asVdc
sin r t + I L
Zr

(3),

vout = Z r Ib sin r t − Vdc cos r t

(4),

ir = I b cos r t +where r is the resonance angle frequency of the resonancecircuit. In order to achieve ZVS of the main device, i.e., todischarge the charges of the parasitic capacitors, vout = VDC
must be satisfied. From (3) and (4), the minimum time td_min toachieve ZVS is expressed as
td _ min =



VDC
1  −1  VDC −1  (5),tan 
 + sin 2
2 2 r 
Z r Ib 
V
+
Z
I

DC
r
b




where Zr is the characteristic impedance of the resonant circuit
and is expressed as

LrCr

The minimum time at which ZVS is achieved in Fig. 2(b)~(d) is also calculated in the same way and agrees with (5).
The parasitic diodes turn on by fully discharging the parasitic
capacitors. Thus, the inductor voltage is clamped to the DC
voltage. As a result, the inductor current is reduced due to achange according to the sign of the inductor voltage. The
parasitic capacitor is recharged by the inductor current
becoming smaller than the load current. In other words, theZVS of the main device fails. The recharging start time td_max
is expressed as

td _ max = td _ min +Lr I b

rVDC

(7).

C. .Selective harmonics elimination
Figure 4 shows a quarter cycle of the p (p: odd) pulse
PWM waveform in a half cycle of an inverter using unipolar
modulation. The switching phase angle bk must be calculatedby the modulation ratio to eliminate the low-order harmonics
components in the PWM waveform. The nth (n: odd) orderharmonic component in the PWM waveform is calculated by
Fourier series expansion of the waveform in Fig. 4 to be
expressed as

an =

p

4VDC 
k +1
 −1 + 2 ( −1) cos nb k 
np 
k =1


= a2 p −1 = 0()

It is impossible to solve these nonlinear simultaneous
equations analytically. Thus, the on/off phase angles bk thateliminate harmonics are solved numerically. Table 1 shows
the on/off phase angles bk when the number of pulses is three,five, and seven, and the fundamental wave amplitude is unity,
respectively.
Figure 5 shows the harmonic analysis results of the SHEPWM waveforms with switching angles in Table 1.
Furthermore, it also shows the harmonic analysis results of thesquare wave used in most conventional WPT systems. The
results of harmonic components are normalized by thefundamental component. From Fig. 5, the square wave
includes many low-order harmonics. The results also showthat the PWM waveform using the numerically analyzed

3 pulse5 pulse7 pulse0.291601942

0.212415864

0.167133468

0.811286668

0.537036587

0.400587541

0.897609093

0.643599531

0.50383472-

1.064356019

0.798263952-

1.093330767

0.847657035--

1.18959039--

1.202551877

0square wave
5-pulse3-pulse7-pulse

-5
-10
-15
-20
-25
-30

1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19
Number of orderFig. 5. Harmonics components in selective harmonics elimination PWM
waveform.
Table2. Simulation conditions.

()

where bk is the switching phase angle of the inverter, there are
p freely variable phases when the number of pulses is p.
Accordingly, p equations are needed to derive all the phaseangles. Since the PWM waveform contains only odd-ordercomponents, all switching phase angles are derived from thefundamental and harmonic components up to 2p−1 order.
Assuming that the amplitude of the fundamental component isunity and the harmonic components are zero, the psimultaneous nonlinear equations are expressed asa1 = 1a3 = a5 =

b1
b2
b3
b4
b5
b6
b7

(6).

PWM output amplitude [dB]Zr =Table 1. Turn-on and turn-off angle for harmonics elimination.

P arameter

SymbolValue

DC voltage

V DC 1, V DC 2

400

Rated output power

P out

1

kW

Transmission frequency

f

85

kHz

V

Self inductance

L 1, L 2

500mH

Resonant capacitor

C 1, C 2

7.02

nF

Coupling coefficientk

0.3

Resonant inductor

Lr

350

nH

P arastic capacitor

Cr

100

pF

Boost current

Ib

6

A

Dead time

td

10.2

ns

Full-bridge inverter output voltage v1 [V]
400
200
0
-200
-400
Diode rectifier output voltage v2 [V]
400
200
0
-200
-400
Primary side coil current i1 [A]
4
2
0
-2
-4
Secondary side coil current i2 [A]
4
2
0
-2
-4
2.5ms/div

Fig. 6. Operation waveform of conventional WPT system.

switching phases has zero fifth order or lower for three pulses,
ninth order or lower for five pulses, and 13th order or lower
for seven pulses. This result confirms the validity of theharmonic analysis of the PWM waveform and the switchingphase angle obtained from the Fourier coefficients.

III. SIMULATION RESULT
Primary ARCP converter output voltage v1 [V]Figure 6 shows the simulation results of a conventional
WPT system with a full-bridge inverter and diode rectifier.
The waveforms show the output voltage and current of the
primary and secondary power converters, respectively. Fig. 6
indicates that the WPT system transmits power under resonantconditions because of the unity power factor. However, it isseen that the current on both the primary and secondary side
are distorted due to the harmonics in the current.

400
200
0
-200
-400

Figure 7 shows the operating waveforms of the proposedsystem, and Table 2 shows the simulation conditions. The

4
2
0
-2
-4

seven-pulse PWM is applied in the simulation, and the on/off
phase angles in Table 1 are used. Fig. 7 shows the primary andsecondary ARCP converter output voltage and thetransmission coil current, respectively. The simulation results
show that the ARCP converter outputs seven pulses in a halfcycle. In addition, the fundamental power factor from theprimary and secondary ARCP outputs is also nearly unity
respectively. In other words, it is seen that the WPT system
operates in the resonance condition.
Figure 8 shows the operating waveforms of the ARCP
converter in the proposed system and the enlarged waveforms
at switching on both the primary and secondary sides. Theyshow the drain-source voltage and the switching signal of the
U-phase upper arm switch, the output current, and the current
in the resonant inductor. The boost current of the proposedsystem is controlled at a constant to minimize the resonant
inductor current while achieving ZVS. From Fig. 8, it is seen

Secondary ARCP converter output voltage v2 [V]
400
200
0
-200
-400
Primary side coil current i1 [A]Secondary side coil current i2 [A]
4
2
0
-2
-4
2.5ms/div

Fig. 7. Operation waveform of proposed WPT system.

that the resonant inductor current changes with the output
current. In addition, the resonant inductor current at the startof the commutation is always 6 A greater than the load current,i.e., the boost current is controlled at a constant level of 6 A.
Due to this, It is confirmed that all switching achieves ZVS.
Figure 9 shows the harmonic analysis results of the
primary and secondary currents, and Table 3 shows the oddorder harmonic components under the 13th harmonic. The
current harmonics when using a two-level inverter are also
shown for comparison. The inverter DC voltage in the two-

Drain-source voltage on S u11 vsu11 [V]Drain-source voltage on S u21 vsu21 [V]

400

400

300

300

200

200

100

100

0

0
Switching s ignal for Su11 [-]Switching s ignal for Su21 [-]

ON

ON

OFF

OFF
ARCP converter output and resonance inductor current i1,ir1 [A]12

ARCP converter output and resonance inductor current i2,ir2 [A]
12

6

i2

6

ir1

0

0

-6

ir2

-6

i1

12

-12
1ms/div

1ms/div
Drain-source voltage on S u12 vsu12 [V]Drain-source voltage on S u11 vsu11 [V]
400

400

300

300

200

200

100

ZVS

0

100

Switching s ignal for Su12 [-]Switching s ignal for Su11 [-]

ON

ON

OFF

OFF

ARCP converter output and resonance inductor current i2,ir2 [A]

ARCP converter output and resonance inductor current i1,ir1 [A]
12

12

-6

6A

i1

6
0

ZVS

0

6

6A

ir2

0

ir1

-6

6A
6A

i2

-12

-12
5ns/di v(a) primary side ARCP converter(b) Secondary side ARCP converter
Fig. 8. Operation waveform of ARCP converter and enlarged waveform at turn-on and turn-off for Su115ns/div

(c) Primary side ARCP converter operation waveform(d) Secondary side ARCP converter operation waveform
Fig. 9. Harmonics analysis for the primary and secondary side coil current in simulation.
Table 3 Analysis results of current harmonics at odd-order components(a) Primary side coil current(b) secondary side coil current

Number of order

Number of order

Configuration

Configuration
1st

3rd

Conv. [dBA]

11.0-16.7 -24.9 -33.4 -35.0 -42.3 -41.1

Prop. [dBA]

11.0-78.5 -90.3 -88.6 -88.3 -97.8 -89.6

5th

7th

9th

11th

1st

3rd

Conv. [dBA]

11.0-16.7 -24.9 -33.4 -35.0 -42.3 -41.1

Prop. [dBA]

11.0-78.5 -90.3 -88.6 -88.3 -97.8 -89.6

13thlevel inverter is multiplied by p/4 so that the fundamental
components of the output voltage are equal. Fig. 9 indicatesthat the fundamental components of the output current are thesame in both systems. In contrast, the harmonic components
are reduced by approximately 40dB from the 3rd to the 13thorder components on both the primary and secondary sides.
Note that the components below the 13th order cannot be
eliminated because the switching phases of the analyticalsolution and the experimental waveform are different due tothe dead time and commutation period of the ARCP converter.
In addition, the current harmonics at uneliminatedcomponents above the 13th order increase. However, thehigher harmonic components are significantly attenuatedaccording to the frequency characteristics of the transmission
coil and the compensation circuit. In other words, they are notexpected to contribute to a significant increase in the radiatedemissions.
IV. EXPERIMENTAL RESULTThe ARCP converter must control the boost current tomatch the switching timing of the main device. However, it
is expected to be more difficult due to the higher resonancefrequency in this paper when the main switch operates at ahigh frequency. Hence, in order to confirm the feasibility of
the ARCP converter, the commutation operation is verified
experimentally, and the fundamental characteristics are
confirmed.
Figure 10 shows the operating waveforms of the WPTsystem with the ARCP converter. Three-pulse PWM isapplied for the experiment, and the DC voltage and theresonance inductance are changed from the simulation
conditions to 100 V and 3 mH, respectively. The waveforms
mention the resonator voltages and transmission coil currentsof the primary and secondary sides, respectively. From theoperating waveforms, the power factor of each power source
is approximately unity. Accordingly, it is confirmed that the
WPT system transmits power in the resonance condition.

5th

7th

9th

11th

13th

Primary coil current i1 [2A/div]
Primary side ARCP converter output voltage v1 [100V/div]0000Secondary coil current i2 [2A/div]
Secondary side ARCP converter output voltage v2 [100V/div][4ms/div]

Fig. 10. Operation waveform of WPT system.

Figure 11 shows the operating waveforms of the primaryand secondary ARCP converters and their enlarged views.
The waveforms are the output voltage of the ARCP converter,
the gate-source voltage and drain current of the U-phase upperarm switch, and the resonant inductor current. From Fig. 8, it
is seen that a negative current flows on the U-phase upper arm
switch Su11 by applying a boost current to the resonant
inductor just before the Su11 turns on. That is, the ZVS of themain switch is confirmed. Note that there is a ringing in theresonant inductor current after the commutation is complete.
The ringing current causes the resonance between the resonant
inductor and the parasitic capacitor of the auxiliary switch
excited by the recovery of the auxiliary switch.
V. CONCLUSION
This paper proposed a configuration of a WPT systemusing an ARCP converter with a synchronous SHE-PWM on
the primary and secondary sides to reduce the coil current
harmonics. The on/off phase angle of the synchronous PWM
was numerically analyzed for seven-pulse PWM. It wasverified by simulation that the proposed system reduces the
harmonics of the coil currents. The results revealed that the
proposed system reduced the 13th-order and lower harmonics
of the transmission coil current harmonics by approximately

Primary side ARCP converter output voltage v1 [100V/div]

Primary side ARCP converter output voltage v1 [100V/div]

0

0

0

0
Gate-source voltage on su11 vgs_u11 [50V/div]

Su1 1 ON

Gate-source voltage on su11 vgs_u11 [50V/div]00Drain current on su1 1 id_u1 1 [2A/div]

Drain current on su1 1 id_u1 1 [2A/div]

Fig. 8. (b)00Primary side resonant inductor current ir1 [5A/div][2ms/div]

Primary side resonant inductor current ir1 [5A/div](a) primary side ARCP converter[40ns/div](b) enlarged waveform of (a)Secondary side ARCP converter output voltage v2 [100V/div]

Secondary side ARCP converter output voltage v2 [100V/div]00Su21 ON00Gate-source voltage on su2 1 vgs_u2 1 [50V/div]

Gate-source voltage on su2 1 vgs_u2 1 [50V/div]

Fig. 8. (d)00Drain current on su2 1 id_u2 1 [2A/div]

Drain current on su2 1 id_u2 1 [2A/div]00Secondary side resonant inductor current ir2 [5A/div][2ms/div](c) Secondary side ARCP converter

[2]

[3]

[4]

[40ns/div](d) enlarged waveform of (c)Fig. 11. Operation waveform of ARCP converter.
[5] International Commission on Non-Ionizing Radiation Protection(ICNIRP): “Guidelines for Limiting Exposure to Electromagnetic
Fields (100 kHz to 300 GHz),” Health Phys., Vol. 118, No. 5, pp. 483524 (2020)[6] N. Misawa： ” Situation of EMC Limit Level of WPT for EV in
CISPR,” 2019 JSAE Annual Congress (Spring) Latest Trend of EV
Charging System, No. 20194438, pp.15-20 (2019)[7] K. Kusaka, J. Itoh: “Development Trends of Inductive Power Transfer
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40 dB. In addition, the commutation operation of the proposedsystem was verified by prototype with the three-pulse SHEPWM. Furthermore, it was confirmed that ZVS was achievedat all switching. In future work, harmonic analysis of coilcurrents and radiated emissions will be measured inexperiments.

[1]

Secondary side resonant inductor current ir2 [5A/div]