Wide Range Multi-level DC-DC Converterwith TCM for High Voltage Applications
Takashi Ohno1, Rintaro Kusui1, Hiroki Watanabe1, Jun-ichi Itoh1, and Takuya Kataoka2
1

Nagaoka University of Technology, Japan2
Mitsubishi Electric Corp., Japan

Abstract— This paper proposes a control method based on
Triangular Current Mode (TCM) for a Current-Fed Dual
Active Bridge (DAB) converter, which uses a three-level
flying capacitor converter in order to accept highvoltage(1500V) output. The proposed control method
achieves the high-efficiency region in a wide operation range.
Additionally, the proposed control performs the Zero
Voltage Switching in a wide operation range. Also, the TCM
is controlled at a variable frequency to keep the negative
current constant. As a result, the operation range is limited
because the switching frequency of the power converter isoften applied to several kilo-Hz in high-power applications.
In order to solve this problem, this paper also proposes thatthe full-bridge and half-bridge operation is adopted using themulti-level flying capacitor configuration. In theexperimental results, the operation range of the proposed
control is extended by 70% compared to that of theconventional Current-Fed DAB converter.
Index Terms—Current-Fed Dual Active Bridge converter,
Multi-level, Triangular Current Mode, Zero Voltage
Switching

I. INTRODUCTION
Recently, medium voltage DC distribution systems
have been widely developed owing to their high systemefficiency and compatibility with renewable energy
sources such as Photovoltaic (PV) and fuel cells [1]-[3]. In
these systems, an isolated DC-DC converter is required tointerface the converter between the DC-bus and energy
sources or batteries.
Several isolated DC-DC converters have been widely
developed for their applications [4]-[8]. A Dual Active
Bridge (DAB) converter is one of the strong candidates
because of its attractive advantages, e.g., 1) simple
configuration, 2) galvanic isolation capability, and 3) Zero
Voltage Switching (ZVS) operation for low switchinglosses. However, the DAB converter has narrow highefficiency characteristics when the voltage transfer ratio
does not match the turn ratio of the high-frequency
transformer. Additionally, the large ripple current may
increase the volume of the filter capacitor.
In contrast, the modulation and modified circuit
configuration of the DAB converter are proposed to
improve the performance. In particular, the modulation
method to expand the ZVS range [9]-[11] and to achieve
the minimum transformer current RMS [12]-[14] havebeen proposed. Specifically, extended-phase shift (EPS)
control, dual-phase shift control, and triple-phase shift(TPS) control have been proposed as a modulation methodthat achieves ZVS over a wide range and a minimum

current RMS. Additionally, the operating frequency of theconverter can also be utilized. With the increase of control
degree of freedom, the challenge for these methods isdifficult to determine the optimal combinations of multiple
control variables, which could include phase shift ratio,
two duty cycles and the operating frequency. As theapproach from the circuit topology, a T-type DAB with theswitching operation mode [15], a six-arm converter withseries-parallel switching [16], and the Neutral Point Clump(NPC) based three-level DAB converter [17] have been
proposed. The T-type topology has improved light load
efficiency during voltage fluctuations. However, T-type is
not enough to achieve high withstand voltage. A six-arm
converter has achieved ZVS over a wide voltage range.
However, efficiency at medium to light loads is very poor.
The NPC-based three-level converter can be achieved highefficiency operation over a wide voltage range and highwithstand voltage. In contrast, the challenge for NPC
converter is many additional components. Additionally,
these converters increase the volume of the filter capacitor
because of the large ripple current.
By contrast, a Current-Fed DAB converter has beenstudied in order to reduce input current ripple or improve
efficiency for several applications [18]-[19]. The CurrentFed DAB converter reduces the ripple current by the
inductor, which is connected to the DC input side. In
particular, the DC capacitor voltage control matches theturn ratio of the high-frequency transformer. In addition,
the Triangular Current Mode (TCM) operation for theCurrent-Fed DAB converter has been proposed [20]. As aresult, the Current-Fed DAB converter achieves ZVS inwide input voltage conditions. However, the switchingfrequency drastically increases at light load conditions
because the TCM operation is implemented based on avariable switching frequency operation. Generally, thehigh-power DC-DC converters for more than several
hundred kilo-watts commonly accept IGBT modules
because the IGBT module has a high voltage and power
rating and is relatively inexpensive. Moreover, theswitching speed of the IGBT module is slower than the SiC
module. As a result, the operating area for the TCM islimited by the switching frequency.
This paper proposes the TCM operation for a CurrentFed DAB converter, which has a three-level Flying
Capacitor (FC) converter on the secondary side, in order toobtain high converter efficiency in light load. The original
points of this paper are two things; the first, the flying
capacitor-based three-level converter is used in thesecondary side and combined by the TCM; the second, thetransformer voltage is changed by switchover the FB and
HB modes depending on the input voltage and thetransmission power. Therefore, the lower voltage rating
devices are used for each switching device. In addition, theFB and HB modes of a three-level converter adjust thetransformer voltage in order to reduce the switchingfrequency of the Current-Fed DAB converter. This paper
is organized as follows; first, the circuit configuration is
introduced. Second, the TCM operating area expansion
method and the operation principle of each operation modeare explained. Third, the efficiency analysis ofconventional and proposed methods is compared. Finally,a prototype of 1500 V output voltage and 12.8-kW output
power is demonstrated by the experiment at several power
conditions in order to confirm the validity of the proposed
control of FB mode and HB mode with TCM.

Secondary Side

Q5

Q9

Q6
VC
iTr N=n1/n2
LDAB
vTr

Q10

Primary Side

Vin

Q1
iLCA

Q3

LCA
iLCB

vPri

LCB
Q2

Q4

VfcA

vSec

Q7

Q11

Q8

Q12

VfcB

Vout

Fig. 1. Circuit configuration of Current-Fed DAB witha three-level flying capacitor converter.
Ton

II. CIRCUIT CONFIGURATION

Q1,Q2

Fig. 1 shows the proposed Current-Fed DAB converterwith a three-level flying capacitor converter. The primaryside converter consists of an H-bridge converter which
integrates the interleaved DC-DC converter. This
converter has the voltage control capability for the DC
capacitor voltage VC to match the voltage ratio with theturn ratio of the transformer. The current mode of iLCA, andiLCB applies the TCM operation to achieve ZVS for the
primary side H-bridge converter. The TCM operation
increases the current ripple compared to the Continuous
Current Mode (CCM) to generate a small negative current.
The interleaved converter reduces the ripple current of the
DC power sources by phase-shifted modulation. A threelevel flying capacitor converter is applied to accept thehigh-voltage secondary DC side. The voltage rating ofeach switching device is reduced to half of the output
voltage Vout.

Q3,Q4
IPeak

iLCB

0
IBottom

iLCA

III. PROPOSED TCM OPERATING AREA EXPANSIONMETHOD

A. Triangular Current Mode Operation
Fig.2 shows the switching pattern of the primary side
H-bridge converter. The switching duty ratio of the typical
DAB converter is set to 0.5. Compared to that, the CurrentFed DAB converter applies the primary DC capacitor
voltage control to adjust the voltage ratio between the input
and output. That is, the switching duty ratio is carried out
depending on the boost ratio of the DC capacitor voltage.
In addition, the variable switching frequency control isimplemented to achieve the TCM operation. As a result,the switching frequency increases at light load conditions
to adjust the negative inductor current. However, theswitching frequency in the TCM is limited by theperformance of the switching devices. As a result, theTCM operation cannot realize in the light load condition.
This paper proposed to use two operation modes, FullBridge (FB) and Half-Bridge (HB) modes on thesecondary side, in order to expand the operating area of the
TCM operation.
The relationship between input power and the switchingfrequency is expressed as

t0

t1

t2

t3

t4

Fig. 2. Switching waveforms with the TCM operation.

Pin =Vin 2 D
+ 2Vin I Bottom ...................................... (1),
LC f S

where IBottom is the boost inductor bottom current, Vin is the
input voltage, VC is the boost capacitor voltage, LC is theboost inductor, fS is the switching frequency, Pin is the
input power, and D is the duty ratio of the lower arm powerdevices on the primary H-bridge, and D is defined as

D = 1−

Vin........................................................... (2).
VC

Namely, the input power is expressed as Vin 
1 −

VC 
V
Pin = in 
+ 2Vin I Bottom ............................ (3),LCfS
2

where N is the turn ratio of the transformer, these are notflexibility parameters because Vin is decided by connected
equipment, VC must match the turn ratio of the transformer,
LC is designed to transmit the rated power with the TCM
operation, and IBottom is decided by the negative current toachieve ZVS. Therefore, the switching frequency has tovary to change the input power. However, the switchingfrequency is limited by power device performance.
According to (3), input power is reduced by reducing
VC, keeping the switching frequency.
B. Full-Bridge Mode
Fig. 3(a) shows the steady-state waveforms of the Full
Bridge (FB) mode of the proposed circuit. According to
Fig.4, the FB operation has eight modes for a switchingcycle.
In addition, only the half-switching cycle (mode I to IV)and  is the phase shift angle between the primary side
voltage and the secondary side voltage. The transmission
power is expressed asof the analysis is enough because the transformer current
is symmetrical. Therefore, the transformer current isexpressed from (4) to (7) as⚫ Mode I ( 0     )VC

iTr _ I ( ) =⚫

 LDAB − iTr _ IV ( ) ............................... (4).

Mode II (     (1 − 2 D) )iTr _ II ( ) =⚫

1 2
vPr i ( )iTr ( )d2 0....................... (8),
VC 2  1 2
=−  + 2 D 
 LDAB  2


PTr _ FB =where vPri is the output voltage of the primary H-bridge.
Note that (4) to (7) are valid when the switching duty D is
0.5 or less. When the switching duty exceeds 0.5, theformula becomes valid by replacing D with (1-D).

VC − NVout( −  ) + iTr _ I ( ) = 0 ......... (5).
 LDAB

Mode III ( (1 − 2 D)    (1 − 2 D) +  )C. Half-Bridge Mode
Fig. 3(b) shows the steady-state waveform of the Half
Bridge (HB) mode of the proposed circuit. According to
fig. 4, the HB operation has eight modes for a switchingcycle. In the HB mode, only a half-switching cycle of theanalysis is needed, as well as in the FB mode.
According to fig. 4, the HB mode applies a half voltageof the output voltage NVout to the primary side of thetransformer. Therefore, the transformer current isexpressed from (9) to (12) as⚫ Mode I ( 0     )VC( − (1 − 2D )  ) + iTr _ II ( (1 − 2D) ) LDAB.............................................................................. (6).

iTr _ III ( ) =⚫

Mode IV ( (1 − 2 D) +      )iTr _ IV ( ) =VC − NVout
 − ( (1 − 2 D) +  )
 LDAB......... (7).

+ iTr _ III ( (1 − 2 D) +  )where Vout is the output voltage,  is the angular frequency,

Q1

Q3

VC

Q5

Q9

Q6

Q10

Q1

Q3

VC

N=n1/n2
LCA

Vin

LCB
Q2

Q4

Q9

Q6

Q10

N=n1/n2

LDAB
vTr

vPri

Q5

LCA

vSe c
Vfc
Q7

Q11

Q8

Q12

Vfc

Vout

Vin

LCB
Q2

LDAB
vTr

vPri

Q4

vSe c
VfcA

VfcB

Q7

Q11

Q8

Q12

(b) Half Bridge (HB) mode(a) Full Bridge (FB) mode

Fig. 3. Switching Pattern of FB and HB mode.
1/fs

NVout
NVout/20-NVout/2-NVoutVout
Vout/20-Vout/2
-Vout

1/fs

0

vPri

vPri

vSec

vSec

iTr

iTr

iLCA

iLCA

vLCA

vLCA

0
Ibottom
Vin0
Vin-Vout/2
Vin-Vout

0 Mode I

II

III

IV



V VI
FB modeVII2 0 
VIII Mode I II IIIIVV VIVII
HB modeFig. 4. Operation waveform of FB and HB modes

VIII

2

Vout

iTr _ I ( ) =⚫

VC − iTr _ IV ( ) ............................... (9), LDAB

Mode II (     (1 − 2 D) )iTr _ II ( ) =⚫

NVout
2 ( −  ) + i ( ) ............ (10),Tr _ I LDABVC −Mode III ( (1 − 2 D)    (1 − 2 D) +  )VC( − (1 − 2D )  ) + iTr _ II ( (1 − 2D) ) LDAB............................................................................ (11),iTr _ III ( ) =⚫

Mode IV ( (1 − 2 D) +      )iTr _ IV ( ) =NVout
2  − ( (1 − 2 D) +  ) .... (12), LDABVC −+ iTr _ III ( (1 − 2 D) +  )Note that VC is halved in voltage to match the voltage with
NVout/2. The transmission power is the same as in the FB
mode, which is expressed as

1 2
vPr i ( )iTr ( )d2 0................... (13),
VC 2  1 2

=
−

+
2

D


 LDAB  2


PTr _ HB =The HB mode applies a half voltage of the output
voltage Vout to the transformer. Therefore, the numerator in(13) becomes smaller; the input power is smaller under thesame operating conditions as the FB mode. In other words,
the TCM operation is achieved when the HB mode isapplied in the low input power range, whereas the TCM
operation is not achieved in the FB mode.
D. ZVS condition
Fig. 5 shows the operating waveform with the ZVS
condition, which is determined by the difference between
the boost inductor current and transformer current. In orderto achieve ZVS, the following condition must be kept
during the dead time period. The ZVS conditions are
expressed as

I

'
LCA

 ITr ........................................................... (14),
'

Vin

'
 I LCA = I bottom + L Td

CA

V
'
C
I = −
Td
 Tr
LDAB
Vin

'
 I LCA = I bottom + L Td

CA

V
V
'
C
I =
 − C Td Tr  LDAB
LDAB

( D  0.5 )( 0.5  D )where ILCA’ is the boost inductor current of A-leg at the endof dead time, LCA is the boost inductor, and Td is the deadtime of the switching device.

According to Fig. 5(a), ZVS is achieved by the TCM
operation under conditions where the switching duty D of
the lower switch is less than 0.5. Because the lower limitof the transformer current is clamped to 0A by the zerocurrent period. However, in Fig. 5(b), the transformer
current flows from zero current to negative current during
the dead time period. Therefore, if the switching duty D of
the lower switch is higher than 0.5, the boost inductor
current must be smaller than the transformer current whenthe end of the dead time. Namely, under the non-ZVS
conditions, the parasitic capacitance of the lower switch isrecharged and hard switching.
E. TCM operating area in FB Mode and HB mode
The TCM operation is implemented using triangularmodulation with the variable carrier frequency. The TCM
operation requires the small negative inductor current todischarge the output capacitance of the IGBT during thedead time before being turned on. The boost inductorbottom current is expressed as

I Bottom =COESVC.................................................. (14),Tdwhere COES is the output capacitance of the IGBT, and Td
is the dead time. IBottom is designed to discharge enoughfrom the parasitic capacitance. Also, the switchingfrequency is increased when the transmission power isdecreased. Thus, the increase in the switching frequency is
reduced by the duty ratio of the lower arm power device
on the primary H-bridge being reduced. That is, the HB
mode of the proposed method is applied to reduce theswitching frequency at the light load conditions so that theTCM operation area is expanded.
Fig. 6 shows the TCM operation area of the FB and HB
modes according to the input voltage and the input current
vGE
0
VC

Td
Q1_vGE

Q2_vGE

Q2_vCE0

iTr

0

IbottomiLCA(a) ZVS operating waveform for 0.5 < D condition

vGE
0
VC
0
0

Td
Q1_vGE

Q2_vGE
Q2_vCE
iTr

IbottomiLCA(b) ZVS operating waveform for D < 0.5 condition
Fig. 5. The relationship between transformer current
and boost inductor current at each duty condition.

(when D < 0.5)
 2 LDABVCVin 2 + 2 LCVC 2Vin f S Td 32
+ LDABVin − LDABVCVin
Pin =.............. (15),LDAB LCVC f S()(when 0.5 < D)

2Vin

( 2L

DAB

Input Current Iin[p.u.]

FB-TCM
Operation area

0.75

LCVC 2Vin 2 + 2 LC 2VC 3Vin ) f S Td

− LDABVin 3 + LDABVCVin 2
LDAB LCVC f S

160.. (16).

IV. EFFICIENCY ANALYSIS
Table 1 shows the efficiency analysis and the experimental
conditions. In this efficiency analysis, only power device
losses are considered, i.e., the magnetic components losses,
and so on, are not considered. The power device losses arecalculated from the switching loss and the conduction loss,
which is expressed as= VCE I C + ES f S

200

240
280
Input Voltage Vin[V]

320

1.0

HB-TCM
Hard Switching atprimary lower arm

0.75

0.5

0.24p.u.

FB and HB modes
ZVS range
160

200

240
280
Input Voltage Vin[V]320Fig. 7. The ZVS range of FB and HB modes

Therefore, the ZVS is achieved when the input power issmaller than (15) or (16). Note that the switching duty ofmore than 0.5 is ZVS in the full operating range because
iTr > iL over the full operating area by applying the TCM.

PLoss = Pcon _ loss + Psw _ loss

0.05p.u.

CCM

Fig. 6. The TCM operation area of FB mode and HBmode according to the input voltage and the input
current

0+ LDAB LCVCVin 3 − LDAB LCVC 2Vin 2

0.44p.u.
HB-TCM
Operation area

0.25
01.0p.u.

Expanded of
TCM operation area

0.5

0.25

+ ( −2 LDABVCVin 2 − 2 LCVC 2Vin ) f S Td
Pin =

1.0

Input Current Iin[p.u.]

obtained by (3). In the FB mode, the TCM operation does
not achieve in the light load condition due to the switchingfrequency limitation at 30 kHz. On the other hand, HB
mode achieves the light load condition from 160 V to 320
V condition. In particular, the 320 V condition is
maximum expanded by 70 % from 0.05 p.u. to 0.44 p.u.
with the operation area. In addition, the ZVS operation isachieved in the light load region below 0.05 p.u. despite
the CCM operation.
Fig. 7 shows the ZVS range of the FB and HB modes.
According to Fig.7, ZVS is achieved in the FB mode overthe full operating range. The hard switching occurs in the
HB mode under conditions where the switching duty is lessthan 0.5. The ZVS range derives the boundary condition
when ITr’ and ILC’ are equal, which is expressed as....................................... (17),where PLoss is the power device losses, Pcon_loss is the
conduction loss, Psw_loss is the switching loss, VCE is the
IGBT’s collector-emitter saturation voltage, IC is thecollector current, and ES is the switching energy. The
collector-emitter saturation voltage is referenced from thedatasheet. The switching loss becomes zero at thecondition achieving an ideal ZVS operation because theenergy of the output capacitance of IGBT is regenerated tothe power supply or sent to the load. However, the otherswitching losses, not caused by the output capacitance,such as the loss caused by a tail current, remain. Therefore,
the energy of the other loss can be obtained by

Table 1. The efficiency analysis and the experimental
conditions.
Element

SymbolValue

Rated power

Prated

12.8kW@Vin=320V

Rated current

Irated40A

Switching frequency

fs(C-F DAB)10-30kHz(DAB) 10kHz

Input voltageVin160-320V

Output voltage

Vout

1500V

Boost capacitor voltage

VC(HB mode) 400V
(FB mode) 800V

Boost inductor

LC

270mH

External inductor

LDAB

45mH

Leakage InductorLl

23.5mH

Magnetizing Inductance

LMg

48mH

Turn ratioN

Dead-time

Td

Hybrid-SiC Modules(C-F DAB) 0.532(DAB) 0.26
1ms
CMH150DY-24NFH

1
ES = Eoff − COESV 2 .......................................... (18),2where Eoff is the turn-off switching energy per pulse fromthe datasheet of the IGBT, and V is the collector-emitter
voltage, which is 750V used in the calculation.
Fig. 8 shows the analyzed efficiency considering only
power device losses. The switching losses dominate at

V. EXPERIMENTAL RESULTS
In this section, the experimental results aredemonstrated in order to evaluate the validity of theCurrent-Fed DAB Converter with a three-level flying
capacitor converter. A 12.8-kW prototype (CMH150DY24NFH, which is Si IGBT + SiC SBD used) is operated at
the switching frequency of 10 kHz to 30 kHz, the input
voltage of 160 V to 320 V and an output voltage of 1500
V.
Fig. 9 shows the operation waveform of the FB mode.
According to Fig. 9(a), the output current of 1p.u., theinput voltage of 320 V, the boost capacitor voltage of 800
V, and the output voltage of 1500 V. Note that theswitching frequency is set to 15.5 kHz. The rated power
operation is confirmed by the experiment. In Fig. 9(b) and(c), the output current is 0.75 p.u., and 0.5 p.u.,
respectively. Also, the phase shift angle between the
primary side voltage and secondary side voltage iscalculated. The phase shift angle of the theoretical value is
9.8 deg. while the experimental value is 8.1 deg. at the
rated power. Meanwhile, the angle of the theoretical valueis 8.7 deg. while the experimental value is 4.9 deg. at the
0.5 p.u. There are errors of 17% and 44% between the

DAB Converter
HB mode(Proposed method)100

Efficiency h[%]

98

97.4%
98.5%

96

FB mode(Proposed method)

94

Conventional C-F DAB(Continuous Current Mode)9290

1p.u.=40A
Vin=320V
0 0.1

0.25

0.75

0.5

Output current Iout[p.u.]

1.0(a) Efficiency analysis of FB mode and HB mode at
input voltage of 320 V.
100

Sec_ConLoss_Diode
Pri_ConLoss_Diode

Loss[W]75

Sec_RecLoss_Diode50

Pri_RecLoss_Diode

Sec_ConLoss_IGBT25

Pri_ConLoss_IGBT
Sec_SWLoss_IGBT0FB

Pri_SWLoss_IGBTHB(b) Efficiency analysis of FB mode and HB mode at
input voltage of 320 V and input current 0.1 p.u.
100
99

Efficiency h[%]light loads due to the high ripple current in the boost
inductor in the Current-Fed DAB converter with the TCM.
Therefore, the light load efficiency is decreased in the FB
mode, where the boost inductor current ripple is larger thanthat in the HB mode. According to Fig. 8(a), the analyzed
efficiency considering only power device losses of the HB
mode is 94 % at the output power of 0.1p.u.. The losses are
reduced by approximately 21.8 % compared to that of the
FB mode. In addition, the conventional method of the
CCM improves the efficiency near the rating by using
TCM because the ZVS is achieved at turn-on by using
TCM. In the light load range, the use of the HB modesignificantly improves efficiency because the HB mode
reduces turn-off losses at light load by reducing the boost
inductor current ripple.
Fig. 8(b) shows the breakdown of losses at the input
voltage of 320 V and the output power of 0.1p.u.. The
analysis results mention the primary side switching loss in
the FB mode is 64 W. In contrast, the primary sideswitching loss in the HB mode is 41 W, which isapproximately 35% less than that of the FB mode. In
addition, the conduction loss is slightly increased byoperating in the HB mode, but it is small enough comparedto the decrease in the switching loss.
According to Fig. 8(c), in the FB mode, the efficiency
analysis at rated current showed that 98% or higher
efficiency was achieved in all voltage conditions and the
maximum efficiency of 98.5 %. On the other hand, losses
increase due to hard switching in the conventional
modulation of the Current-Fed DAB. Although, the DAB
converter had the maximum efficiency in Fig. 8(a). The
efficiency drops significantly due to voltage fluctuations,especially in low-voltage conditions.
Fig. 8(a) clearly shows that the HB mode is unsuitable
for the rated current operation. For this reason, Fig. 8(c)does not show the efficiency of the HB mode.

FB mode(Proposed method)

98.5%

98.0%

98

Const. Iout=40A

97

DAB Converter
96
95
160

Conventional C-F DAB(Continuous Current Mode)
200

240

Input voltage[V]

280

320

(c) Efficiency analysis of FB mode at rated output
current conditions.
Fig. 8. Efficiency analysis of FB mode and HB mode
at input voltage 320 V.
angle of the experimental waveform and the theoreticalvalue, respectively. The cause of this phase shift angle
error is the output capacitance of the IGBT and operating
conditions error where the load current is small. The error
compensation will be studied in the future.
Fig. 10 shows the operation waveform of the HB mode.
According to Fig. 10(a), the output current of 0.43 p.u., theinput voltage of 320 V, the boost capacitor voltage of 400
V, and the output voltage of 1500 V. Note that theswitching frequency is set to 10 kHz. In Fig. 10(b), theoutput current of 0.3 p.u. and in Fig. 10(c), the output
current of 0.2 p.u.. In the FB mode, the switchingfrequency of 30kHz is required to obtain the input powerof 0.43 p.u.. The switching frequency required in the HB

mode is lower than that in the FB mode at light loads. The
switching frequency is reduced by 66% at 0.43p.u. using
the HB mode. According to Fig. 9 and Fig. 10, the negative
current is kept constant by variable frequency control
depending on the load, that is achieved the TCM operation.
Fig. 11 shows the efficiency characteristics with each
operation mode at Vin = 320 V and Vout = 1500 V. The
maximum efficiency of the experimental results for the FB
and HB mode is 3.7% lower than the efficiency analysis
because other losses except the power device are notconsidered, such as the transformer, and the inductor
losses. The transformer current is increasing in the HB
mode compared to the FB mode by the boost voltage of VC
is half. Therefore, the efficiency of the HB mode is higher
than the FB mode at the light load where turn-off loss
dominates the whole loss. In contrast, the efficiency of the
FB mode is higher than the HB mode from the medium torated loads where conduction loss dominates the wholeloss. Fig. 11, the maximum efficiency of 94.8 % in the FB

mode is achieved at the rated power. Also, the maximum
efficiency of the HB mode is 93.7 % at the 43 % input
current. Furthermore, the HB mode has a voltage polarity
inversion during the dead time period. The polarity
inversion occurs when the transformer current is equal tothe boost inductor current. Also, the primary lower switch
is hard-switching because of the voltage polarity inversion.
More negative current of boost inductor is needed toachieve the ZVS, but the optimal inductor current should
be considered in the future due to increased conduction
loss.
VI. CONCLUSIONSThis paper proposed a three-level Flying Capacitor
Current-Fed DAB converter for high output voltage
applications(1500VDC). The proposed converter achieved

 = 8.08deg.

vPri [1kV/div]

vPri [500V/div] Voltage polarity inversion

00

vSe c [1kV/div]vSe c [2kV/div]00

iTr [20A/div]

iTr [20A/div]00iLCA [40A/div]0

iLCA [20A/div]

48.0A

-3.2A
Negative current period

20[µsec/div]0

 = 6.25deg.

vPri [1kV/div]

30[µsec/div]

vPri [500V/div]

Voltage polarity inversion

00

vSe c [1kV/div]  = 20.9deg.

vSe c [2kV/div]00

iTr [20A/div]

iTr [20A/div]00iLCA [40A/div]

iLCA [20A/div]

37.1A

-2.8A
Negative current period

20[µsec/div]030[µsec/div]

vPri [500V/div]Voltage polarity inversion

 = 4.91deg.

vPri [1kV/div]

Negative current period(b) Vin = 320V, Iin = 0.3p.u., fS = 12.8kHz(b) Vin = 320V, Iin = 0.75p.u., fS = 19.8kHz00vSe c [1kV/div]vSe c [2kV/div]

 = 18.5deg.

00

iTr [20A/div]

iTr [20A/div]00iLCA [20A/div]

iLCA [40A/div]
25.6A
0Negative current period(a) Vin = 320V, Iin = 0.43p.u., fS = 10.0kHz(a) Vin = 320V, Iin = 1.0p.u., fS = 15.5kHz0 = 18.8deg.

Negative current period-3.2A

10[µsec/div](c) Vin = 320 V, Iin = 0.5 p.u., fS = 27.4kHz
Fig. 9 FB mode operation waveform of each input
current conditions.

0

Negative current period

20[µsec/div](c) Vin = 320 V, Iin = 0.2 p.u., fS = 16.9kHz
Fig. 10 HB mode operation waveform of each input
current conditions.

94.8%

95

HB modeEfficiency h[%]

94

93.7%
Vin=320V
1p.u.=40A

93

FB mode
92

91.8%

91
90

0

0.25

0.5

0.75

Output current Iout[p.u.]

1.0

Fig. 11. Efficiency characteristic of FB mode and HBmode at input voltage of 320 V.
the TCM operation area over the wide transmission power
range by the FB mode and the HB mode. In particular, it
was confirmed that with the application of the HB mode,
the TCM operation area was extended to 70 %. The
validity of the proposed method was demonstrated by a
12.8-kW prototype circuit with Silicon IGBTs.
The maximum efficiency was 94.8 % of the FB mode at
rated power conditions. On the other hand, the maximum
efficiency was 93.7 % of the HB mode at the input voltageof 320V and the input current of 0.44 p.u. From theexperiment, the TCM was achieved with a constantnegative current by the variable switching frequency
control. The light load efficiency was also improved by the
HB mode compared to the FB mode from the efficiency
analysis result and the experiment. The future works are
the following, to obtain efficiency characteristics at othervoltages and, load conditions; loss separation including
iron loss; the compensation method for transmission
power error, caused by the IGBT’s output capacitance.
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