Application Report
SCAA082A – November 2006 – Revised August 2017

High-Speed Layout Guidelines
Alexander Weiler, Alexander Pakosta, and Ankur Verma ........................................................ Clock Drivers
ABSTRACTThis application report addresses high-speed signals, such as clock signals and their routing, and gives
designers a review of the important coherences. With some simple rules, electromagnetic interference
problems can be minimized without using complicated formulas and expensive simulation tools. Section 1
gives a short introduction to theory, while Section 2 focuses on practical PCB design rules. Either section
can be read independently.

1

2

3
4

Contents
Theoretical Overview ........................................................................................................ 1
1.1
Electromagnetic Interference and Electromagnetic Compatibility ............................................ 1
1.2
Clock Signals ........................................................................................................ 2
1.3
Transmission Lines ................................................................................................. 3
1.4
Crosstalk ............................................................................................................. 8
1.5
Differential Signals .................................................................................................. 8
1.6
Return Current and Loop Areas .................................................................................. 8
Practical PCB Design Rules ................................................................................................ 9
2.1
PCB Considerations During the Circuit Design ................................................................. 9
2.2
Board Stackup....................................................................................................... 9
2.3
Power and Ground Planes ....................................................................................... 10
2.4
Decoupling Capacitors............................................................................................ 13
2.5
Traces, Vias, and Other PCB Components .................................................................... 14
2.6
Clock Distribution .................................................................................................. 17
Summary .................................................................................................................... 19
References .................................................................................................................. 19

Trademarks
All trademarks are the property of their respective owners.

1

Theoretical Overview
Some basic understanding is desirable to effectively use the PCB design rules given in this document. It isthen easy to identify the undesirable effects that can arise and how to avoid them. The reason PCB layout
becomes more and more important is because of the trend to faster, higher integrated, smaller form
factors, and lower power electronic circuits. The higher the switching frequencies are, the more radiation
occurs on a PCB. With good layout, many EMI problems can be minimized to meet the required
specifications.

1.1

Electromagnetic Interference and Electromagnetic Compatibility
Electromagnetic interference (EMI) is radio frequency energy that interferes with the operation of anelectronic device. This radio frequency energy can be produced by the device itself or by other devices
nearby.
Electromagnetic compatibility (EMC) is the ability of an electronic product to operate without causing EMI
that would interfere with other equipment and without being affected by EMI from other equipment or theenvironment.

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The goal is to reduce EMI to meet the requirements given by the Federal Communication Commission(FCC) or the International Special Committee on Radio Interference (CISPR)
A basic EMI model is shown in Figure 1. Every device acts as a source and simultaneously as a sink. It
can cause interference through a coupling path and can be affected by interference through the coupling
path. The coupling can be:• Capacitive• Inductive• Galvanic• Radiated power
SourceSink
Coupling Path
SourceSink

Figure 1. Model of Electromagnetic Interference
There is not just one coupling mechanism present, but rather a combination of them. With proper PCB
layout, however, these effects can be reduced.

1.2

Clock Signals
Figure 2 illustrates the time and the frequency domain of a clock signal. Ideally, it is a square wave., but inreality, it is not possible to change from low level to high level (and vice versa) in an infinite short time.
Due to the rise and fall time, it has the shape of a trapezoid in the time domain. By means of the Fourier
series, the trapezoid consists of a series of sine and cosine signals with different frequency andmagnitude. The discrete frequency components have an envelope as is shown in the lower diagram of
Figure 2. An important aspect is that in the frequency domain the amplitude of the higher frequency
harmonics depends on the rise and fall time of the signal. The longer the rise time, the smaller themagnitude of the harmonics. For example, the harmonics of a 100-MHz clock signal are not negligible,
especially the third and fifth. In this case, consideration also should be made with frequencies up to 500
MHz. With the CDCE906 from Texas Instruments, the user can set different rise and fall times to reduce
the amplitude of the harmonics. However, take care to ensure these times do not violate the slew rate
specifications of the driven devices.

2

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Figure 2. Time Domain and Frequency Domain of a Clock Signal

1.3

Transmission Lines
If the lengths of traces are in the range of the signal's wavelength, then the user has to consider theeffects of transmission lines. The problems that a user must deal with are time delay, reflections, andcrosstalk. To get a better understanding of these problems and where and how they arise, it is useful toknow what transmission lines are. They are simply the traces on a PCB and depend on the length and thefrequency of the signals passing through them.
Many different structures of trace routing are possible on a PCB. Two common structures are shown in
Figure 3. On the left, a microstrip structure is illustrated, and on the right, a stripline technique. A
microstrip has one reference, often a ground plane, and these are separated by a dielectric. A stripline has
two references, often multiple ground planes, and are surrounded with the dielectric.

Figure 3. Structure and Dimension of Microstrip and Stripline
The following sections describe some important properties of transmission lines which are significant for
PCB design. Many software tools are available to calculate the properties of the several transmission line
structures. In this application report, the freeware AppCAD from Agilent is used so that the reader canbecome familiar with these properties. Figure 4 shows the two structures, microstrip (top) and stripline(bottom). The dimensions, the material, and the frequency are in each case shown on the left. The resultsthat are used in the following sections can be seen on the right. Table 1 defines the symbols used in
Figure 3 and Figure 4.

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Table 1. Description of the Symbols Used in Figures 3 and 4
SYMBOLDESCRIPTIONH

Height of the dielectric

W

Width of the trace

L

Length of the trace

Frequency

The frequency on which the calculations are based

Z0

Characteristic impedance of the trace

1.0 Wavelength

Wavelength λ of the trace at the given frequency and the given effective dielectric
l+

4

3

10 8 mńs
Ǹå
ƒ
eff

VP

Velocity of the signal on this trace with the given dimensions and frequency relative to the speed of light. The
absolute velocity is calculated by VP,.absolute = VP,.relative × 3 × 108 m/s

εeff

Combination of the several dielectrics which surrounds the microstrip

W/H

Ratio between trace width and trace length

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Figure 4. Calculation of Properties of Microstrip and Stripline (AppCAD)

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Signal Speed and Propagation Delay TimeA signal cannot pass through a trace with infinite speed. The maximum speed is the speed of light with 3
× 108 m/s. For a certain trace length, the signal needs a certain time to pass it, and this is called the
propagation delay time. The standard medium for the speed of light is air. For another medium, thedielectric in a PCB environment, the speed is different than that of the speed of light in air. The formula forthe speed on a stripline is:
3 108 mńs
V+
Ǹå r

(1)

So, the speed is a function of the dielectric which surrounds the trace. For a microstrip, it is more
complicated because the trace is not surrounded by one dielectric. There are at least two: the substrateunder the trace and the air above the trace. If the PCB contains a solder mask, a third medium would bepresent. Therefore, the calculation of an effective εr is necessary before determining the signal speed. It
depends on the width of the microstrip and the distance to the reference plane. In this case, the speed is afunction of the present dielectric, the trace width, and its distance to the reference plane [12].
The signal speed and the propagation delay time, respectively, on a signal trace are important when:• Timing and skew requirements must be met (clock distribution, buses, and so forth)• Differential traces were used (for example, LVDS)
1.3.1.1

ExamplesThe following parameters are the same in each case:length = 100 mm; thickness = 35 μm; height = 1.5 mm; εr = 4.6 (FR4); frequency = 300 MHz
In Table 2, the dependency of signal speed on the trace width at the microstrip structure is shown. Using a
stripline, the signal speed is not a function of the trace width. The same is valid for the height.
Table 2. Comparison of Propagation Delay Time

(1)

1.3.2

WIDTH
[mm]

εr,eff

VP,relative

VP,absolute[mm/ns]Pd (1)[ps/100 mm]

Microstrip 1

0.5

3.046

0.573

171.9

581.7

Microstrip 2

1

3.165

0.562

168.6

593.1

Stripline 1

0.5

4.6

0.466

139.8

715.3

Stripline 2

1

4.6

0.466

139.8

715.3

Pd is the propagation delay time in ps on the mentioned line with A 100-mm length.
1
Pd +100 mm171.9 mmńns

Characteristic Impedance, Reflections, and Termination
Another property of a transmission line is the characteristic impedance, Z0. The microstrip in Figure 4 hasfor the given attributes a characteristic impedance Z0 = 105 Ω, and the stripline Z0 = 55 Ω. If there are any
impedance changes in the signal chain (source – trace – vias – connectors – sink, and so forth),reflections occur. These reflections cause over- and undershoots. The two extreme are a transmission linewith an open end (R = ∞) and a shorted end (R = 0 Ω). The reflection coefficient ρ is the dimension which
expresses the relationship between the impedance of the transmission line and the impedance of theR * Z0ò+.
R ) Z 0 For the two mentioned cases, the reflection coefficient
source or sink. It is calculated as
becomes ρ = +1 for an open end and ρ = –1 for a shorted end. The value 1 means that the complete
signal reflects at this impedance change and goes back to the source. To avoid this, the reflection
coefficient must be ρ = 0 to get no reflections. This is the case if the impedance at the source has thesame value as the characteristic impedance of the transmission line.

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In Figure 5, the case with an open end (high-impedance input stage of the sink) is simulated. The clock
source has an output of 3.3 V and an impedance of 25 Ω. The red line (dotted) is the ideal shape of theclock output. The green one (dashed) is the real signal at the clock's output and the blue line is at the endof the transmission line. The reflection is the cause of these over- and undershoots. The maximum voltage
at the trace end is approximately 4.4 V instead of 3.3 V, and the minimum voltage is approximately –1 V
instead of 0 V. These circumstances can damage the input stage of the source and sink.

Figure 5. Over- and Undershoots Due to Incorrect Termination
A receiver often has a high-impedance input. To avoid these over- and undershoots, the reflections must
be reduced. Therefore, a proper termination is required. The most common termination techniques follow:• Series termination• Parallel termination• Thevenin termination• AC termination

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Each of them has advantages and disadvantages, and the designer has to trade off which one is the bestsolution for his design.
References: [8], [9]1.4Crosstalk
The mutual influence of two parallel, nearby routed traces is called crosstalk. One is called the aggressor(this trace carries the signal) and the other is called the victim (this trace is influenced by the aggressor).
Due to the electromagnetic field, the victim is influenced by an inductive and a capacitive coupling. They
generate a forward and a backward current in the victim trace whereas in a homogenous environment (forexample, stripline) the two induced currents cancel each other. In a microstrip environment, the forward
current of the inductive coupling tends to be larger than the influence of the capacitive coupling. To
minimize the effects on crosstalk on adjacent traces, keep them at least 2 times the trace width apart.
Transmission lines coupled through a media have crosstalk associated with them. There are two types ofcrosstalk: forward crosstalk and backward crosstalk. Forward crosstalk travels in same direction as theaggressor. It grows with length and increasing aggressor dV/dt, whereas the backward crosstalk travels in
the opposite direction as the aggressor and depends only on the aggressor amplitude. Make sure tomeasure the crosstalk between channels v/s frequency. IEEE802.3KR specifies Insertion Loss Crosstalk
Ratio (ICR) that sums the crosstalk and plot it using S21.
References: [1], [2], [3], [4], [5]1.5Differential Signals
In the case of differential signals, the negative effect of crosstalk is a positive one. The signals on eachtrace are in theory exactly equal in magnitude and opposite in their sign. So, their electromagnetic fields
cancel each other and the return current on the ground plane, as well. To achieve this, the traces for bothsignals must have the same length so that the propagation delay times are equal. The receiver is sensitive
to the differences of the signals and not to an absolutely level reference, for example, to ground. If any
signal radiated equally into both differential traces, the receiver would not recognize it. Therefore, thedesigner has to make sure that radiation affects both traces equally. This can be realized by routing thedifferential traces as close as possible.
References: [1], [6], [7]1.6

Return Current and Loop Areas
An electrical circuit must always be a closed loop. Up to now, only the signal path was discussed but not
the path back to the source – the return current. With DC, the return current takes the way back with thelowest resistance (Figure 6a). With a higher frequency, the return current flows along the lowest
impedance (Figure 6b). This is directly beside the signal.

Figure 6. Return Current and Resulting Loop Area
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If this return path, mostly the ground plane, has a slot, the return current has to take another way and thisresults in a loop area (Figure 6c). The larger the area, the more radiation and EMI problems occur. The
designer has to make sure that the return current can flow directly underneath the signal trace. One
solution is to place a 0-Ω resistor over the slot (Figure 6d). Another is to route the signal the same way as
the return current flows. The best solution is to avoid any slots in the ground reference plane.

2

Practical PCB Design Rules
Because many things can affect transmission lines, EMI problems can occur. To reduce these problems,
good PCB design is important, and with some simple design rules, the PCB designer can minimize these
problems. It is important to make prudent decisions during new circuit design, like the minimum number oflayers. The easiest way to get a good, new design is to copy the recommended design from the TI
evaluation modules (EVM).
A good PCB layout starts with the circuit design. Do not postpone considerations about the layout. One of
the most important aspects affecting the layout is the location of each functional block. Keep their devices
and their traces together.

2.1

PCB Considerations During the Circuit Design
•
•
•
•
•
•
•
•

What is the highest frequency and fastest rise time in the system?
What are the electrical specifications at the inputs and outputs of the sinks and sources?
Are there sensitive signals to route – for example, think about controlled impedance, termination,
propagation delay on a trace (clock distribution, buses, etc.)?
Is a microstrip adequate for the sensitive signals, or is it essential to use stripline technique?
How many different supply voltages exist? Does each supply voltage need its own power plane, or is it
possible to split them?
Create a diagram with the functional groups of the system – for example, transmitter path, receiverpath, analog signals, digital signals, and so forth
Are there any interconnections between at least two independent functional groups? Take special careof them. Think about the return current and crosstalk to other traces.
Clarify the minimum width, separation and height of a trace with the PCB manufacturer. What's theminimum distance between two layers? What about the minimum drill and the requirements of vias? Is
it possible to use blind vias and buried vias?

Equipped with this information, a designer can do a lot of basic design.

2.2

Board Stackup
There is no fundamental information about how many layers should be used and how the board stackup
should look. Again, the easiest way the get good results is to use the design from the EVMs of Texas
Instruments. The magazine Elektronik Praxis [11] has published an article with an analysis of different
board stackups. These are listed in Table 3 and Table 4.
Generally, the use of microstrip traces needs at least two layers, whereas one of them must be a GND
plane. Better is the use of a four-layer PCB, with a GND and a VCC plane and two signal layers. If thecircuit is complex and signals must be routed as stripline, because of propagation delay and/or
characteristic impedance, a six-layer stackup should be used.
Table 3. Possible Board Stackup on a Four-Layer PCB
Model 1Model 2Model 3Model 4
GND

Layer 1

SIG

SIG

SIG

Layer 2

SIG

GND

GND

SIG

Layer 3

VCC

VCC

SIG

VCC

Layer 4

GND

SIG

VCC

SIG

Decoupling

Good

Good

Bad

Bad

EMC

Bad

Bad

Bad

Bad

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Table 3. Possible Board Stackup on a Four-Layer PCB (continued)
Model 1Model 2Model 3Model 4Signal integrity

Bad

Bad

Good

Bad

Self disturbanceSatisfaction

Satisfaction

Satisfaction

High

Table 4. Possible Board Stackup on a Six-Layer PCB
Model 1Model 2Model 3Model 4Model 5Model 6

SIG

SIG

GND

SIG

SIG

SIG

Layer 2

SIG

GND

VCC

GND

GND

GND

Layer 3

VCC

VCC

SIG

VCC

VCC

VCC

Layer 4

GND

VCC

SIG

SIG

GND

GND

Layer 5

SIG

GND

VCC

GND

Not used

SIG

Layer 1

Layer 6

SIG

SIG

GND

SIG

SIG

SIG

Decoupling

Good

Good

Good

Good

Good

Good

EMC

Bad

Good

Satisfaction

Satisfaction

Good

Good

Signal integrity

Bad

Good

Bad

Good

Good

Bad

To determine the right board stackup, consider the following points:• Define the location of each section on the board by means of the functional diagram. Try to keep thedevices together to avoid interaction (crosstalk, influence of noise) between two separate blocks (liketransmitter–receiver or analog–digital).
• At which functional block is which supply voltage used?
• It is necessary in high-speed designs to have at least one complete ground plane as a reference formicrostrip traces for sensitive signals.
• With a complete power plane as close as possible to the ground plane, it is possible to create
capacitive coupling between them to get low impedance at high frequencies. This reduces the amounton small decoupling capacitors at the power pins of the devices. The closer the planes, the less
impedance is present [13].

2.3Power and Ground Planes
As previously mentioned, a complete ground plane in high-speed design is essential. Additionally, acomplete power plane is recommended as well. In a complex system, several regulated voltages can bepresent. The best solution is for every voltage to have its own layer and its own ground plane. But thiswould result in a huge number of layers just for ground and supply voltages. What are the alternatives?
Split the ground planes and the power planes? In a mixed-signal design, for example, using data
converters, the manufacturer often recommends splitting the analog ground and the digital ground to avoid
noise coupling between the digital part and the sensitive analog part.
Take care when using split ground planes because:• Split ground planes act as slot antennas and radiate.
• A routed trace over a gap creates large loop areas, because the return current cannot flow beside thesignal, and the signal can induce noise into the nonrelated reference plane (Figure 7).
• With a proper signal routing, crosstalk also can arise in the return current path due to discontinuities in
the ground plane. Always take care of the return current (Figure 10).
Figure 7: Do not route a signal referenced to digital ground over analog ground and vice versa. The returncurrent cannot take the direct way along the signal trace and so a loop area occurs. Furthermore, thesignal induces noise, due to crosstalk (dotted red line) into the analog ground plane.

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Figure 7. Loop Area and Crosstalk Due to Poor Signal Routing and Ground Splitting
Figure 8: Do not let one ground plane pass another ground plane to get connected to the common ground(a). Every ground plane must have its own path to the common ground to reduce noise (b).

Figure 8. Poor and Good Placement of the Common Ground in a Split Ground Environment
Figure 9: The use of a complete ground plane is a better solution. Place the devices by function and route
their signals only in their region. If any interconnection between analog and digital occurs, be careful withcrosstalk and the return current path.

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Figure 9. Good Placement of Different Functional Blocks Without the Need of a Split Ground Plane
Figure 10: If the trace routing of two signals is done properly, it is still possible to induce noise in the returncurrent path by means of crosstalk. In this case, a loop area and crosstalk in the return current path havebeen created.
• If possible, use a continuous ground plane; do not split them. This can be achieved by a proper
placement selection. Again, create functional blocks, and place and route them together. By doing this,the traces of a digital part cannot influence any trace of the analog part if these sections do not crosseach other.
If split ground planes are essential:• Do not route signals over a gap. Always strive for the return current flow with the smallest loop area.
• Connect split ground planes only at one point. More common ground connections can create groundloops, and this increases radiation.
• The return current of a subsystem (for example, an analog system or transmitter path) must not be in
the path of the other subsystem (digital system or receiver path). The return current should flow directly
to the common ground point (Figure 8).
• Power planes should only reference their own ground plane. They should not overlap with anotherground plane. This leads to capacitive coupling between the power plane and a not-referenced groundplane. Noise can couple into the other system.
• Do not connect bypass capacitors between a power plane and an unrelated ground plane. Again,
noise can be coupled from one supply system into the other. This mistake can occur in the circuitdesign section.

Figure 10. Crosstalk Induced by the Return Current Path

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2.4

Decoupling Capacitors
Decoupling capacitors between the power pin and ground pin of the device ensure low AC impedance toreduce noise and to store energy. To reach low impedance over a wide frequency range, several
capacitors must be used. This is why, a real capacitor consists of its capacitance and a parasitic
inductance and resistance. Therefore, every real capacitor behaves as a resonant circuit. The capacitive
characteristics are only valid up to its self-resonant frequency (SRF). Above the SRF, the parasitic effects
dominate, and the capacitor acts as an inductor. With the use of several capacitors with different values,
low AC impedance over a wide frequency range can be provided.

Figure 11. Impedance of Different Capacitors Over a Wide Frequency Range and the Resulting
Impedance of Their Parallel Connection
Figure 11 shows this context. Capacitors with high values have low impedance in the lower frequency
range and a low SRF, whereas small-valued capacitors have their SRF in the upper frequency range. Thisdepends on the equivalent series resistance (ESR) and the equivalent series inductance (ESL). A goodcombination of several capacitors leads to a low impedance over a wide frequency range. This is shownwith the Cparallel curve in Figure 11. The gap (increase of the impedance) at around 60 MHz is the resultof a missing capacitance. If there were a value between 100 nF and 10 nF, the Cparallel curve would notincrease.
As previously mentioned, a power and GND plane can represent a capacitance that ensures low
impedance at high frequencies. Therefore, a well-designed board stackup can minimize the number ofcapacitors required having low-capacitance values.
General rules for placing capacitors:• Place the lowest valued capacitor as close as possible to the device to minimize the inductive influenceof the trace. This is especially important for small capacitor values, because the inductive influence ofthe trace is not negligible anymore.
• Place the lowest valued capacitor as close as possible to the power pin/power trace of the device.
• Connect the pad of the capacitor directly with a via to the ground plane. Use two or three vias to get a
low-impedance connection to ground. If the distance to the ground pin of the device is short enough,you can connect it directly.
• Make sure that the signal must flow along the capacitor.

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Poor Bypassing

Good Bypassing

Figure 12. Poor and Good Placement and Routing of Bypass Capacitors
The decoupling or the bypass capacitance must provide the high frequency components of the current.
For example for Advanced Driver Assistance Systems (ADAS), the camera current fluctuates between
active and blanking periods. Hence, bypass capacitors must take care of the ripple due to non-uniform
current within a clock cycle, handle load transients, and filter any noise that could cause the high speed
signal to be corrupted. Parallel capacitors are used to give wideband low impedance.

2.5

Traces, Vias, and Other PCB Components
A right angle in a trace can cause more radiation. The capacitance increases in the region of the corner,
and the characteristic impedance changes. This impedance change causes reflections.
• Avoid right-angle bends in a trace and try to route them at least with two 45° corners. To minimize any
impedance change, the best routing would be a round bend (see Figure 13).
• Separate high-speed signals (for example, clock signals) from low-speed signals and digital fromanalog signals; again, placement is important.
• To minimize crosstalk not only between two signals on one layer but also between adjacent layers,route them with 90° to each other.

Figure 13. Poor and Good Right Angle Bends
The use of vias is essential in most routings, but the designer has to be careful when using them. They
add additional inductance and capacitance, and reflections occur due to the change in the characteristic
impedance. Vias also increase the trace length.
• Avoid vias in differential traces. If it is impossible to avoid them, use vias in both traces or compensate
the delay also in the other trace.

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Figure 14. Loop Areas Caused by Poor Via Placement
Figure 14 illustrates another problem with vias – loop areas. The designer has to make sure that thereturn current can flow ideally underneath (beside) the signal trace. A good way to realize this is to add
some ground vias around the signal via. This is a similar structure to a coaxial line. Take care if the newlayer has another distance to the reference plane; this can cause reflections due to an impedance change.
Figure 15 shows how good via placement avoids loop areas if there is need for multiple vias.

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Figure 15. Poor and Good Via Placement
Figure 15: Because of wrong via placement, especially with bus signals, slots in the ground plane canarise. Further, the return current can create a loop area when placing the vias in the wrong direction.
Three different routings are shown on the left. On the right, some traces are masked out to see how the
return current (green trace) will flow.

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In Figure 16, a four-layer stackup can be seen. The signal changes from layer 1 to layer 2. The via passes
through all four layers. Thus, we get the structure of an open-end transmission line with no terminationand thereby a 100% reflection. Due to the length of the stub, a delay also occurs. The worst case is now
that the two signals (original one → red and reflected one → gray) on layer 2 have a phase shift of 180°,and so they cancel each other. The phase shift is due to the delay which arises from the stub length. The
time delay is twice the stub length divided by the signal speed. To avoid this problem, use blind vias or
buried vias.

Figure 16. Reflection Caused by Stubs in a Via
Tips for routing traces and the use of vias:• Do not use right-angle bends on traces with controlled impedance and fast rise time, respectively.
• Route the traces orthogonally to each other on adjacent layers to avoid coupling.
• To minimize crosstalk, the distance between two traces should be approximately 2 to 3 times the widthof the trace.
• Differential traces should be routed as close as possible to get a high coupling factor. As a result of
this, influenced noise is then a common-mode noise and is not a problem for a differential input stage.
• Do not use vias on traces with sensitive signals, if unnecessary.
• Be careful with the return current when changing the layers. Use ground vias around the signal via tomake sure that the return current can flow as close as possible to the signal (see Figure 14).
• Do not create slots, for example in the ground plane, by using closely placed vias (see Figure 15).
• Consider stubs created by vias. If necessary, use blind vias or buried vias (see Figure 16).
Micro strip implementation for traces can have high crosstalk and are not recommended for high-speed
digital systems. Stripline traces inherently have zero crosstalk and minimize the EMI/EMC (see An
EMC/EMI system-design and testing methodology for FPD-Link III SerDes (SLYT719) for more
information). Striplines require vias because traces have to buried inside the PCB. Vias do offer someparasitic inductance in the path, however, good news is that the impedance of a properly designed via canbe controlled to avoid any reflections on the line. Before concluding that vias are good, consider anotherscenario in which a PCB via is passing through completely through the printed-circuit board. If the traceconnections are made at one end of the via that leaves a stub on the other side. This stub would look likea capacitor on the TDR plot and can create resonance phenomena. At the quarter wave frequency, theopen-ended stub looks like a short and produces nulls of transmission. Thus, the designer must back drill
such vias after the PCB is fabricated. The dangling via stub can be trimmed by drilling from the back side.

2.6

Clock Distribution
Figure 17 shows four possible clock distribution circuits. The problems in Figure 17a are the enormousreflections at the branches and the different trace length to the devices. Because of the delay, it ispossible that the system cannot function properly. To avoid the reflection at the branches, do not use them(see Figure 17b). Route the signal in a chain from one device to the other and realize a proper
termination. Be careful with the delay. In a high-speed environment, it is possible that the data, sent fromdevice A to device B, is out of date when the clock signal arrives at device B. A star connection as shown

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Practical PCB Design Rules

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in Figure 17c is a good solution to minimize the delay. A clock driver is used to distribute the clock signals
to the different devices. To minimize the skew, the same trace length for every clock signal should be
used. Figure 17d illustrates a solution for a complex system. A main clock feeds several clock drivers.
Again, to reach low skew, implement delay-time compensation and use a proper termination to avoidreflections.
• Be careful with trace length in a clock distribution layout. Consider the delay for each trace. The bestsolution is to route these signals with the same length.
• Avoid branches to reduce reflections. Use a clock driver to distribute the signal to every device andconsider a proper termination.

Clock
Driver

Clock

a)

c)

Clock
DriverClock
DriverImpedance
Matching

A

Clock
Generator

Clock
DriverClock
DriverMatching ofLine Length

data-line

Clock
Driver

B

1 nF

Clock

100 W

b)

d)

Figure 17. Poor and Good Clock Distribution on a PCB

18

High-Speed Layout Guidelines

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Summary

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3

SummaryThis document presents an introduction to designing a PCB, a complex topic. However, the rules
presented in this introduction can assist the designer in creating proper PCB designs. The higher thesignal frequency with which the designer must contend, the more complicated will be the PCB design.
Complicated PCB designs require a deep knowledge, experience, and simulation tools. However, it is notalways necessary to route traces as short as possible, differential signals as close as possible, or to avoid
crosstalk as much as possible. Rather, it depends on the signal on the trace. Basically, the designer must
know which are the sensitive parts in the circuit or where problems due to reflections can occur. With thisknowledge, a good placement of the devices can be made. Because placement is such an important stepin high-speed design, the designer will do well to always keep it and the return current in mind.

4

References
1. High-Speed DSP Systems Design Reference Guide (SPRU889)
2. Crosstalk, Part 1: The conversion we wish would stop, Douglas Brooks,
http://www.ultracad.com/articles/crosstlk.pdf
3. Crosstalk, Part 2: How loud is your crosstalk?, Douglas Brooks,
http://www.ultracad.com/articles/crosstlk.pdf
4. Crosstalk, EMI and differential Z, Douglas Brooks,
http://www.ultracad.com/articles/crossemianddifferentialz.pdf
5. Crosstalk, Part 1: Understanding Forward vs Backward, Douglas Brooks,
http://www.ultracad.com/mentor/mentor%20crosstalk%20part1.pdf
6. Differential Impedance: What's the difference, Douglas Brooks,
http://www.ultracad.com/articles/diff_z.pdf
7. Differential Signals, Douglas Brooks, http://www.ultracad.com/articles/differentialrules.pdf
8. Adjusting signal timing (part 1), Douglas Brooks,
http://www.ultracad.com/mentor/mentor%20signal%20timing1.pdf
9. Transmission line termination, Douglas Brooks,
http://www.ultracad.com/mentor/transmission%20line%20terminations.pdf
10. EMV-Design Richtlinien, B.Foeste/s.Oeing, ISBN: 3-7723-5499-8, Franzis
11. Elektronik-Praxis: Die Leiterplatte 2010, Ausgabe 11/06, P72-77
12. Microstrip propagation times – slower than we think, Douglas Brooks,
http://www.ultracad.com/mentor/microstrip%20propagation.pdf
13. Design for minimizing electromagnetic interference in high frequency RF and digital boards andsystems, Dr. Eric Bogatin, www.bethesignal.com
14. Ankur Verma, An EMC/EMI system-design and testing methodology for FPD-Link III SerDes(SLYT719), 2Q’ 2017, Analog Applications Journal

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Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (November 2006) to A Revision ................................................................................................ Page••

20

Added content to Crosstalk section ..................................................................................................... 8
Added content to Decoupling Capacitors section ................................................................................... 13

Revision History

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