T-type NPC Inverter with Active Power Decoupling Capability using
Discontinuous Current Mode
Akiyoshi Omomo, Nagisa Takaoka, Hoai Nam Le, Keisuke Kusaka, Jun-ichi Itoh
Nagaoka University of Technology
1603-1 Kamitomioka-cho
Nagaoka city Niigata, Japan
Tel.: +81 / (258) – 47.9533.
E-Mail: omomo@stn.nagaokaut.ac.jp, ntakaoka@stn.nagaokaut.ac.jp,
lehoainam@stn.nagaokaut.ac.jp, kusaka@vos.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp
URL: http://itohserver01.nagaokaut.ac.jp/itohlab/index.html

Keywords«DC power supply», «Single phase system», «Microgrid»

Abstract
This paper proposes a power decoupling control method based on a discontinuous current mode fora T-type NPC inverter. In general, additional components such as buffer capacitors and inductors are
required in order to provide a power decoupling circuit to inverters. These components lead an increasein circuit volume and weight. In the proposed current control method, the power decoupling capability
is achieved without any additional components. Furthermore, the grid-tied inductor, which is connectedto the output side, is minimized because the discontinuous current mode (DCM) is used for the current
control. As a simulation result, the second order harmonic component of the input current is reduced by
90.2% compared to that without power decoupling control. Then, the proposed power decoupling
control is confirmed with a T-type NPC inverter as a prototype. Consequently, the second orderharmonic component of the input current is reduced by 77.1% compared to that without power
decoupling control, whereas the output current THD is 3.13% at 200-W load.

I. Introduction
In recent years, DC micro grid and DC power supply systems have been actively studied [1–7].
However, most of the household electronic equipment still use only AC input voltage. Therefore, theDC-AC converter is needed to be connected as an interface between the DC grid and the AC equipment.
When connecting the DC grid to the single-phase AC grid, a power decoupling stage is usually requiredin the converter. The reason is that the input of the DC-AC converter is the DC power, whereas theoutput power of the DC-AC converter oscillates at twice the grid frequency. As a result, a power ripple
at twice the grid frequency occurs at the DC-link of the DC-AC converter. In order to absorb this power
ripple, bulky electrolytic capacitors are generally used for the power decoupling capability. However,
the electrolytic capacitor limits the lifetime of the system [8–9].
As an alternative power decoupling method, active power decoupling methods, which use buffer
capacitors, inductors, and switching devices in order to eliminate the use of the electrolytic capacitor,
have been proposed [10–13]. In general, film capacitors or ceramic capacitors are used as the buffer
capacitors, so the converter with a long lifetime is achieved. However, additional components for the
power decoupling circuit lead to an increase in circuit volume and weight.
In order to avoid the additional components, the power decoupling method using the T-type NPC
inverter is proposed [14]. This method uses only the grid-tied inductor to carry out the control of thegrid current to the sinusoidal waveform, and the control of the neutral point current, which helps toachieve the power decoupling method, at the same time. Note that the neutral point current is defined as
the current, which flows into the neutral point. In the conventional method, the continuous current mode
(CCM) is employed, so the two current controls have an interference with each other. Because theflexibility of current control decreases, either the output current or the neutral point current is impossible
to be controlled. The period which those current are not simultaneously controlled is called “current
uncontrollable period” in this paper.
In this paper, the novel active power decoupling method for T-type NPC inverter is proposed inorder to eliminate the interference between the decoupling control and the grid current control. In
particular, CCM is used instead of DCM to independently control two currents at the same time without
additional magnetic component. Furthermore, the grid-tied inductor can be minimized by the use of
DCM. The contribution of this paper is that both the elimination of the electrolytic capacitors and theminimization of the system can be achieved simultaneously. This paper is organized as follows; first,
the principle of the active power decoupling method is explained, second, the circuit configuration andthe conventional active power decoupling control for T-type NPC inverter is explained. Then, theproposed active power decoupling control based on DCM is introduced.
As the simulation result, the current ripple of the input current is reduced by 90.2% compared to thatwithout the power decoupling control, whereas the output current THD is 1.87%.
Finally, as the experimental result, the current ripple of the input current is reduced by 77.1%
compared to that without the power decoupling control, whereas the output current THD is 3.13%.

II. Principal of Active Power Decoupling Method
In this chapter, the reason why the power ripple occurs in a DC-AC circuit is explained. The output
voltage and the output current of the inverter are respectively presented byvout = 2Vout sin ( ωt )(1),

iout = 2Iout sin (ωt )(2),where Vout is the RMS value of vout, Iout is the RMS value of iout, and ω is the grid angular frequency.
Thus, the instantaneous output power is expressed bypout = vout iout = Vout I out − Vout I out cos(2ωt )(3),= Pconst + p fluctwhere Pconst is the constant value and pfluct is the fluctuation value. Equation (3) shows that the inverter
produces a power ripple at twice the grid frequency whereas the input power is constant.
Figure 1 shows the relationship among the input power, the instantaneous output power, and thebuffer power. The DC input power should be constant, whereas the instantaneous output power oscillates
at twice the grid frequency. In order to absorb the power ripple, the difference between the instantaneous
input power and the instantaneous output power should be compensated by the charge and discharge ofthe buffer capacitor as shown in Fig. 1(b). The relationship of the compensation power and the charge
and discharge energy of the capacitor is given by1
ΔE = C (Vmax 2 − Vmin 2 )(4),2
where ΔE is the buffer energy, which is absorbed by the capacitor, C is the capacitance, Vmax and Vmin
are the maximum and minimum value of the oscillated voltage of the buffer capacitor. As shown in (4),the compensation energy is proportional to the capacitance. In the conventional method, a large
electrolytic capacitor is connected between the DC grid and the DC-AC converter in order to absorb thepower ripple.
pout =Pin =Vm I m2

Fig. 1. Single-phase power ripple compensation.

Vm I m(1 − cos 2ωoutt )2

On the other hand, in the active power decoupling circuit, the additional switching devices,
inductors, small capacitors are used for the active buffer circuit instead of adopting a bulky electrolytic
capacitor in the DC-link [10–13]. The voltage of the buffer capacitor is oscillated by the control of thecurrent on the additional inductor. As a result, the buffer capacitor absorbs the same power ripple witha small capacitance due to the increase of voltage ripple Vmax2-Vmin2. Thus, a film or ceramic capacitors
can be used instead of the electrolytic capacitor to increase the lifetime of the system.

III. Circuit Configuration and Conventional Method
Figure 2 shows the circuit configuration of the single-phase T-type NPC inverter. The T-type NPC
inverter is possible to operate in both the directions. In this paper, the power flow from the DC side to
the AC side is considered. In the conventional method in [14], the active power decoupling control isachieved by the oscillation of the voltages of two capacitors C1 and C2.
Figure 3 shows the waveforms of the DC-link voltage and the two capacitor voltages. The twocapacitor voltages VC1 and VC2 are given by
V
VC1 = dc + Vm sin(ωt + δ )(5),2
VVC 2 = dc − Vm sin(ωt + δ )(6),2
where Vdc is the DC-link voltage, Vm is an amplitude of the capacitor voltage, ω is the angular gridfrequency, and δ is the phase difference to the output voltage. As shown in Fig. 3 and (5–6), because
the power decoupling control is achieved by oscillating the two capacitor voltages in opposite phases,
the DC-link voltage is the sum of the two capacitor voltages, which is constant. From (5–6), the capacitor
currents iC1 and iC2 are represented as
dV
iC1 = C C1 = CωVm cos(ωt + δ )
(7),
dt
dV
iC 2 = C C 2 = −CωVm cos(ωt + δ )(8),dt
where C is the capacitance of the buffer capacitor C1 and C2. From Kirchhoff’s law, neutral point current
is represented as
in = iC1 − iC 2 = 2CωVm cos(ωt + δ )(9),Fig. 2. DC to single-phase AC T-type NPC inverter.

t

Fig. 3. Waveforms of ideal capacitor voltage and DC-link voltage. The two capacitor voltages oscillate
in opposite phases.

From (5–9), capacitor voltages are controlled by flowing neutral point current. In addition, from (5–8),
the instantaneous capacitor power is represented as
1
pC = CωVm 2 sin(ωt + δ )cos(ωt + δ ) = CωVm 2 sin(2ωt + 2δ )(10).
2In order to absorb the power ripple, (10) has to be same as pfluct in (3). In (10) and pfluct in (3), thevariation is only sinusoidal function term. Thus, the constant value and the variation can be compared
respectively. The constant value, which means amplitude, and the variation, which indicates phase, arerespectively compared in (11–12)− cos(2ωt ) = sin(2ωt + 2δ )
(11)
2
(12).
Vout I out = C ωVm
From (11), the initial phase difference of neutral point current δ is determined. Furthermore, from(12), the amplitude of the capacitor voltage Vm is determined. Then, Vm is expressed by
V I
(13).
Vm = out outCωNote that Vm should not exceed the half of Vdc because the DC-link voltage is the sum of the twocapacitor voltages,
VVm ≤ dc(14).
2
According to (9) and (13), the neutral point current which flows to the neutral point is expressed by
π

(15).
in = 2 Vout I out Cω cos  ωt − 4Through the control of the neutral point current, the capacitor voltages are oscillated as (5–6) toabsorb the power ripple.
Figure 4 shows the relationship of the absolute value of the current commands between the gridcurrent and the neutral point current. If the single-phase T-type NPC inverter employs with CCM as in
the conventional method [14], the neutral point current control and the output current control areinterfered with each other during the period where the command value relationship becomes |in*|>|iout*|
as shown in the shaped area in Fig. 4. When the neutral point current flows during that period, the current
which is larger than the output current command value, flows to the grid-tied inductor. Thus, the output
current is affected by the neutral point current. In the conventional method, only the output current is
controlled in the period in order to prevent interference. As a result, the power ripple remains in the DClink since the neutral point current is different from command value.

IV. Proposed Active Power Decoupling Method
In the proposed method, DCM is employed to the current controls of the single-phase T-type NPC
inverter drawn in Fig. 2. In DCM, there is zero-current interval, where the inductor current becomeszero and all the switching devices are turned off. For this reason, the interference of the two current
control is solved by flowing each current during the zero-current interval of the other current. In

Fig. 4. Absolute current command value of output current and neutral point current. In the shaped areaof Fig. 4. The neutral point current control and the output current control are interfered with each otherin the conventional CCM method.

particular, the neutral point current flows during the zero-current interval of the grid current and viceversa. If the zero-current interval of each current is designed to be long enough to flow the other current,
two currents can be independently controlled at the same time.
Figures 5 and 6 show the current waveform of the neutral point current and the output current of the
T-type NPC inverter drawn in Fig. 2 with DCM. As shown in Fig. 5, the neutral point current in flows
in DCM, resulting in the zero-current interval.
By the control of this neutral point current according to (15), the capacitor voltages oscillateaccording to (5–6), achieving the power decoupling control. As shown in Fig. 6, the inductor current iL
flows during the zero-current interval of the neutral point current so that the sum of iL and in results inthe sinusoidal grid current. Thus, two currents are simultaneously controlled using only grid-tied
inductors, leading to the achievement of the power decoupling control and the sinusoidal grid current
control at the same time.
Figures 7 and 8 show the switching patterns, which are used for the control of the neutral point
current and the output current. The neutral point current is controlled using a grid-tied inductor as shownin Fig. 7(a–d). In addition, the output current is controlled after the neutral point current becomes zero.
Note that the output current does not flow through the current path of the neutral point current.
Figure 9 indicates the proposed control block diagram for the T-type NPC inverter with the power
decoupling operation. The upper block diagram of Fig. 9 controls the output current and the output
voltage. Note that in_pol* is a polarity of the neutral point current which flows to the grid-tied inductor.
The lower block diagram of Fig. 9 controls the average voltage of the capacitor and the neutral point
current. Therefore, the average voltage of the capacitor is controlled to the half value of the DC-link
voltage with the PI controller. Note that the cutoff frequency of the PI controller is configured low to
avoid the interference with output voltage PI controller. Meanwhile, the capacitor voltage is oscillatedby the feed forward of iL*, which follows (7). Then, the duty ratios are calculated from the current
commands to generate the switching signals. As these results, the proposed method with DCM achieves
a compensation of the remaining power ripple on DC-link and controlling the output current.
Figure 10 (a) shows the relationship between the ideal compared capacitor voltage VC_large* and theideal absolute value of the output voltage |vout*|. Note that the capacitor voltage indicates only the larger
voltage of VC1 and VC2. Since the circuit is buck type, the VC_large* must be higher than |vout*| during theoutput period. For this reason, the maximum voltage |vout*| is Vdc 2 , the maximum RMS value of vout*is Vdc / 2. In this paper, the modulation rate is defined as 1.0 when the configured RMS value of vout* is
Vdc / 2.

Fig. 5. Waveform of in operated in the DCM

Fig. 6. Waveform of iL operated in the DCM

Figure 10 (b) shows the relationship between the ratio of the RMS value of in and iout and themodulation rate. Note that C / Cmin is defined as 100% when the Vm of (13) is Vdc / 2. In the proposed
method, the sum of the neutral point current in and the output current iout flows to the inductor and theswitching devices. Thus, the conduction loss increases as in increases. As shown in Fig. 10 (b), the ratio,
In_rms / Iout_rms decreases as the modulation ratio decreases or C / Cmin decreases. As a result, the circulating
current is low in the applications requiring low modulation ratio and low capacitance.

(a) in increases, iL increases.

(b) in increases, iL decreases.

(c) in decreases, iL increases.
(d) in decreases, iL decreases.
Fig. 7. Switching patterns of neutral point current in control. in and iL are controlled simultaneously.

(b) iL decreases.
(a) iL increases.
Fig. 8. Switching patterns of output current iL control. Only iL flows by using those switchingpatterns whereas in is zero.
iL

*
vout

vout

*

d1

in

VDC
2

VC2

d2

*

in_polpolarity (vin )
*
in
*

d3
iL

d4

*

Fig. 9. Proposed control block diagram for T-type NPC inverter.
C= 200%
Cmin

Vdc
2

Vdc
2

(a) Relation of VC_large and |vout*|(b) Relation of current ratio and modulation rate
Fig. 10. Relation of voltages, current ratio and modulation rate.

V. Simulation Results
Table I shows the simulation conditions, whereas Fig. 11(a) and (b) show the waveforms of the input
current, the capacitor voltages, the output voltage, the output current without/with the proposed power
decoupling capability respectively.
As shown in Fig. 11(a), the input current oscillates at twice the grid frequency because the use of
the small capacitance of C leads to the high current ripple on DC-link according to (3). On the other
hand, as shown in Fig. 11(b), the current ripple of the input current is reduced significantly compared tothat in Fig. 11(a) because the capacitor voltages VC1, VC2 oscillate to absorb the power ripple in the
proposed power decoupling method.
Figure 12 shows the harmonic analysis of the input current. By the proposed method, the componentof 100 Hz, i.e., twice the grid frequency is reduced by 90.2% compared to that result without the power
decoupling method whereas the THD of the output current is 1.87%.

Idc[A]
VC1,VC2[V]
iout[A]
iL,in[A]vout[V]iout[A]
iL,in[A]vout[V]

VC1,VC2[V]

Idc[A]

Table I: Simulation parameters(a) Operation without power decoupling control.
(b) Operation with power decoupling control.
Fig. 11. Waveforms of input current, capacitor voltages, output voltage and output current.in
steady state.

Fig. 12. Harmonic component of input current.

VI. Experimental Results
Table II shows the experimental parameters. The experiments have been done with an open-loop. In
general, percent impedance %ZL of grid-tied inductor is designed around 5%. However, as shown in
Table II, the grid-tied inductor of the prototype is 50μH (the two inductors 25μH are connected in series).
Hence, the %ZL is 0.16%. Furthermore, the additional inductor which is required for conventional power
decoupling control is not needed.
Figure 13 (a) and (b) show the waveforms of the output voltage, the output current, and the input
current with/without the proposed power decoupling method at 200-W load respectively. From Fig. 13(a), the output voltage is controlled in sinusoidal waveform at 50 Hz whereas the input current iscompensated by the proposed method. The THD of the output current is 3.13%.
Figure 14 (a) and (b) show the waveforms of the two capacitor voltages, the output current, and theinductor current (DCM current) with/without the proposed power decoupling method at 200-W load
respectively. Note that the output current is republished to confirm the phase relationship between

Table II: System parameters(a)Operation waveforms (w/ power decoupling) (b)Operation waveforms (w/o power decoupling)
Fig. 13. Measured output voltage, output current and input current. Due to the proposed DCM
current control method, the power decoupling control is achieved whereas the output current THD
is 3.13% at 200W load.

(a)Operation waveforms (w/ power decoupling) (b)Operation waveforms (w/o power decoupling)
Fig. 14. Measured capacitor voltage, output current and inductor current. Due to the proposed
DCM current control method, the two capacitor voltages are fluctuated in opposite phases.

fluctuated two capacitor voltages and the output current.
From Fig. 14 (a), the two capacitor voltages are fluctuated in opposite phase by controlling neutral
point current. From (2) and (5), the theoretical phase difference between the fluctuation component ofcapacitor voltage and the output current is π/4. However, the real phase difference is smaller than thetheoretical phase difference. The reason is that the increases of the power loss due to the circulation
current by applying the power decoupling control. In the theoretical consideration, the power loss of thecapacitor and the inductor is assumed as zero. However, the power loss cannot be ignored in theexperiment. Thus, the conditions of the neutral point current command is changed.
Figure 15 shows the enlarged view of the inductor current. From Fig. 15, the output current and theneutral point current are controlled simultaneously in one switching period.
Figure 16 shows the harmonic analysis of the input current. By applying the proposed method, thecomponent of 100 Hz, i.e., twice the grid frequency is reduced by 77.1% compared to that result without
the power decoupling control.
Figure 17 shows the fluctuated capacitor voltages and the relationship of the current command values|in*| and |iout*|. As shown in Fig. 4, the power decoupling control cannot be applied when |in*| is higher
than |iout*| in the conventional method. Because of the current uncotrollable period, the two capacitor
voltages are distrorted. However, from Fig. 17. the fluctuation conponents of the capacitor voltage are

Fig. 15. Measured enlarged view of inductor current. Output current and neutral point current arecontrolled simultaneously in one switching period.

Fig. 16. Harmonic component of input current.

Fig. 17. Measured capacitor voltages and absolute value of current commands. Power decoupling
control is achieved when |in*| is higher than |iout*|controlled in sinusoidal waveform when |in*| is higher than |iout*|, i.e., the proposed power decoupling
method can be applied in all time.
From (13) and Table II, the ideal amplitude of the two capacitor voltages is calculated as 61.9 V.
From Fig. 17, the amplitude of the two capacitors is mesured as 63.8 V. Note that the amplitude iscalculated by half of peak to peak of the capacitor voltage. The Error rate of the capacitor voltage
amplitude is 3.08%. Thus, the proposed control method is employed correctly.

VII. Conclusion
This paper proposed the power decoupling method based on DCM for T-type NPC inverter without
any additional passive components. The proposed control method achieved the low ripple current at theDC-link and the low grid current THD. Thus, the reduction of the system size and the long lifetime are
expected from the proposed method. In particular, the current ripple of the input current at twice the gridfrequency is reduced by 90.2%. Moreover, the output current THD is 1.87%.
In the proposed method, the ratio of the neutral point current to the output current decreases with thedecrease of the modulation rate and DC-link capacitor. Therefore, the circulating current is low in thecase of applications requiring low modulation rate and low capacitance.
As the experimental result, the two capacitor voltages are fluctuated by flowing the neutral point
current. The output voltage and power decoupling control are achieved without additional component.
As a result, the second harmonics component of the input current is reduced by 77.1% compared to that
result without the power decoupling control whereas the THD of output current is 3.13%.
In the future work, applying feed-back control, operation at rated load 1-kW, and formulation of theneutral point current command which include the influence of the power loss due to the circulation
current will be considered in order to improve the efficiency.

References[1] Y. Ito, Z.Yang, and H. Akagi, “DC micro-grid based distribution power generation system”, IEEE Int. PowerElectron. Motion Control Conf., Vol. 3, pp. 1740–1745 (2004)[2] X. Lu, J. M. Guerrero, K. Sun, “An Improved Droop Control Method for DC Microgrids Based on Low
Bandwidth Communication With DC Bus Voltage Restoration and Enhanced Current Sharing Accuracy”, IEEE.
Trans Vol. 29, pp. 1800-1812 (2014)[3] T. Dragičević, J. M. Guerrero, J. C. Vasquez, “Supervisory Control of an Adaptive-Droop Regulated DC
Microgrid With Battery Management Capability”, IEEE. Trans, Vol. 29, pp. 695-706 (2014)[4] X. Lu, K sun, J. M. Guerrero, J. C. Vasquez, “State-of-Charge Balance Using Adaptive Droop Control forDistributed Energy Storage Systems in DC Microgrid Applications”, IEEE. Trans, Vol. 61, pp. 2804-2815 (2014)[5] V. Nasirian, S. Moayedi, A. Davoudi, F. L. Lewis, “Distributed Cooperative Control of DC Microgrids”, IEEE.
Trans Vol. 30, pp. 2288-2303 (2015)[6] Toru Tanaka, Yoshinori Takahashi, Kenji Natori, and Yukihiko Sato”High-Efficiency Floating Bidirectional
Power Flow Controller for Next-Generation DC Power Network,” IEEJ J. Industry Applications, vol. 7, no. 1, pp.
29-34, (2018)[7] Toshiki Nakanishi, and Jun-ichi Itoh,”Control Strategy for Modular Multilevel Converter based on Singlephase Power Factor Correction Converter”,IEEJ J. Industry Applications, vol.6, no.1, pp.46-57, (2017)[8] H. Hu, S. Harb, X. Fang, D. Zhang, Q. Zhang : "A Three-port Flyback for PV Microinverter Applications With
Power Pulsation Decoupling Capability", IEEE. Trans, Vol. 27, No. 9, pp. 3953-3964 (2012)[9] Kodai Abe, Hitoshi Haga, Kiyoshi Ohishi, and Yuki Yokokura,”Current Ripple Suppression Control Based on
Prediction of Resonance Cancellation Voltage for Electrolytic-Capacitor-Less Inverter”,IEEJ J. Industry
Applications, vol.6, no.1, pp.1-11, (2017)[10] T. Sakuraba, J. Itoh, H. N. Le, K. Kusaka, “Requirements for Circuit Components of Single-Phase Inverter
Applied with Power Decoupling Capability toward High Power Density”, Power Electronics and Applications
2016, (2016)[11] H. Hu, S. Harb, N. Kutkut, I. Batarseh, Z. J. Shen : “Power Decoupling Techniques for Micro-inverters in PV
Systems — a Review,” Energy Conversion Congress and Exposition 2010, pp.3235-3240, pp. 12-16 (2010)[12] K. H. Chao, P. T. Cheng: "Power Decoupling Methods for single-phase three-poles AC/DC converters",
Energy Conversion Congress and Exposition 2009, pp. 3742-3747 (2009)[13] S.Yamaguchi, T.Shimizu: “Single-phase Power Conditioner with a Buck-boost-type Power Decoupling
Circuit”, IEEJ Journal of Industry Applications, Vol. 5, No. 3, pp. 191-198 (2016)[14] M. Abe, H. Haga, S. Kondo, “Electrolytic capacitor-less single-phase AC/DC converter using T-type NPC
circuit”, Telecommunications Energy Conference 2015, (2015)