T-type NPC Inverter with Active Power Decoupling
Method using Discontinuous Current Mode forMicro-Inverter
Akiyoshi Omomo, Jun-ichi Itoh, Keisuke Kusaka, Nagisa Takaoka, Hoai Nam Le*Nagaoka University of Technology
Nagaoka, Niigata, Japan
itoh@vos.nagaokaut.ac.jp

Abstract—This paper presents a power decoupling control
method for a T-type NPC inverter with a discontinuous current
mode (DCM). The proposed method does not require additional
components for active power decoupling method. Furthermore,
the grid-tied inductor is minimized because the output current of
the inverter is in DCM. The operation of the proposed method is
confirmed with a prototype. Consequently, the second orderharmonic component of the input current is reduced by 67.7%
compared to that without the power decoupling control, whereas
the output current THD is 1.9% at an output power of 500 W.
Keywords—Micro inverter; PV system; Active power decoupling;
Discontinuous current mode.

I. INTRODUCTION
Solar energy is attracting the attention as one of the dominant
renewable energy sources [1–5]. In recent years, micro inverters
are actively employed for photovoltaic systems in order toensure maximal power from photovoltaic panels [6–10]. Whenthe micro inverters are connected to AC grids, a power
decoupling capability is required in the DC side. The reason isthat there is an instantaneous power difference between the input
and the output of a DC-AC inverter. In order to suppress thepower ripple, in general, large electrolytic capacitors are
installed to absorb the power ripple at double line frequency.
However, the electrolytic capacitors have high losses due to theirhigh ESR, suffer from short life time compared with film
capacitors [11].
As the power decoupling method, the power decoupling
controls with an additional circuit using film or ceramic
capacitors, inductors and switching devices, have been alreadyproposed [12–14]. System life time becomes longer by
employing ceramic or film capacitors instead of the electrolytic
capacitors. However, its cost and weight increase because of theadditional circuit for power decoupling control.
In order to avoid applying additional components, the power
decoupling method for single phase T-type NPC inverter is
proposed [15]. This method simultaneously achieves an output
current control and a power decoupling control despite one
inductor. Note that the current for achieving power decoupling
control is defined as “neutral point current” in this paper because
the power decoupling control is achieved by flowing the current
through the neutral point in the T-type NPC inverter. In theconventional method, Continuous Current Mode (CCM) is
applied for the current control method. In general, one inductor
is required for one current control in CCM. On the other hands,
the circuit has only one inductor. Thus, there are the terms thatthe output current control and neutral point current control
interrupt each other. The term is named “current uncontrollable
period” in this paper.
In this paper, the active power decoupling method for single
phase T-type NPC inverter in order to eliminate the current
control interruption. As the feature of the proposed method,
DCM is applied for the current control method instead of CCM
in order to avoid the current control interruption without
additional inductors. Moreover, the output side inductor isminimized by applying DCM. The advantage of the proposed
method is that both elimination of the electrolytic capacitors andthe minimization of the system can be simultaneously achieved.
This paper is organized as follows; first, the principle of theactive power decoupling method is explained, second, the circuit
configuration and the conventional active power decoupling
control for T-type NPC inverter is explained. Third, theproposed active power decoupling control based on DCM is
introduced. Finally, the operation of the proposed method is
confirmed in the simulation and the experiment at an output
power of 500 W.
II. PRINCIPLE OF ACTIVE POWER DECOUPLING METHOD
In this section, the principle of an occurring power ripple ina DC-AC circuit and absorbing the power ripple by a decoupling
control is explained. The output voltage and the output currentof an inverter are respectively represented asvout  2Vout sin t (1),iout  2 I out sin t (2),where the capital letter indicates the RMS value of theirinstantaneous value, and  is the grid angular frequency. Thus,
the instantaneous output power is represented as

pout  vout iout  Vout I out  Vout I out cos(2t ) PActive  pRe active(3),

where PActive is the active power which is a constant value
and pReacct is the reactive power, which is fluctuation value. Eq.
3 indicates that a power ripple at twice the grid frequency is
occurred in an inverter whereas the input power is constant.
Figure 1 shows the relationship among the input power, theoutput power, and the compensation power in the buffer
capacitor. From Fig. 1, the output power oscillates at twice thegrid frequency, whereas the input power should be constant. In
order to compensate the power ripple, the instantaneous power
difference between the input and the output is compensated bycharging and discharging the buffer capacitor according topReactive in Eq. 3 as shown in Fig. 1(b). The relationship of thepReactive and the charged and discharged energy of the buffer
capacitor is represented as1
E  C Vmax 2  Vmin 2 
2

pin

Pin 

Vm I m2

Constant value0

where E is the buffer energy, which is the decoupling
capability of the buffer capacitor, C is the capacitance, Vmax isthe maximum and Vmin is the minimum value of the oscillated
voltage of the buffer capacitor. As shown in Eq. 4, thecompensation energy is dependent to the capacitance and thevoltage ripple of the capacitor. In general, bulky electrolytic
capacitors are used at the DC-Link stage in DC-AC converter inorder to absorb the power ripple.
However, in the active power decoupling method, additionalswitching devices, inductors, small capacitors are employed forthe active buffer circuit instead of applying bulky electrolytic
capacitors in the DC-link [12–14]. The buffer capacitor voltage
is oscillated by controlling the current on the inductor in the
power decoupling circuit. As a result, the buffer capacitor
absorbs the same power ripple with a small capacitance due tothe increase of voltage ripple Vmax2-Vmin2 in Eq. 4. Therefore, afilm or ceramic capacitors can be used instead of the electrolytic
capacitor to increase the lifetime of the system.
III. CIRCUIT CONFIGURATION AND PROPOSED ACTIVE POWER
DECOUPLING MEHOTD
Figure 2 shows the circuit configuration of the single-phase
T-type NPC inverter. In T-type NPC inverter, the power flow isbidirectional. In this paper, the power flow is defined from theDC side to the AC side. In the conventional method in [14], theactive power decoupling control is achieved by the oscillation of
the two capacitor voltages VC1 and VC2 in opposite phase.
Figure 3 shows the ideal waveforms of the two capacitor
voltages and the summation of the two capacitor voltages which
is DC-Link voltage. The two capacitor voltages VC1 and VC2 arerepresented as

VC1 

Vdc
 Vm sin(t   )2(5),

VC 2 

Vdc
 Vm sin(t   )2(6),

where Vdc is the DC-link voltage, Vm is an amplitude of the twoof capacitor voltages,  is the angular grid frequency, and  isthe initial phase against the output voltage. As shown in Fig. 3=

discharge

Vm I m1  cos 2outt 
2

+

0

t

0

t

t
Tout

charge(b)Buffer power(a)Input power(c)Output power

Fig. 1. Principal of power ripple compensation in Single-phase DC-AC
converter.

Lfilter
C1
iC1

(4),

pout 

pbuf

Vdc

S1

S2

S5,6

iL
L1

in

vout

iC2
Lfilter

iout

L2
C2

S4

S7,8

S3

Fig. 2. Single phase T-type NPC inverter. The output current control andneutral point current are controlled by only one series of inducotor L1 and
L2

VC
Vdc

Vdc: constantVm

VC1

VC_aveVC2t
Fig. 3. Waveforms of ideal capacitor voltages and the summation of thetwo capacitor voltages which is DC-link voltage. The two capacitor
voltages are oscillated in opposite phase.

and Eqs. (5–6), the DC-link voltage is the summation of the twocapacitor voltages, which is constant, because the power
decoupling control is achieved by oscillating the two capacitor
voltages in opposite phases. From Eqs. (5–6), the capacitor
currents iC1 and iC2 are given byiC1  C

dVC1
 CVm cos(t   )
dt

(7),

iC 2  C

dVC 2
 CVm cos(t   )
dt

(8),

where C is the capacitance of the two of DC-Link capacitors C1
and C2. The neutral point current is calculated asin  iC1  iC 2  2CVm cos(t   )(9),From Eqs. (5–9), capacitor voltages are controlled by theneutral point current. In addition, from Eqs. (5–8), theinstantaneous capacitor power is given bypC  CVm 2 sin(t   )cos(t   )1
 CVm 2 sin(2t  2 )
2

(10).

In order to absorb the power ripple, Eq. 10 has to be same as
pReacitve in Eq. 3. In Eq. 10 and pReactive in Eq. 3, the variation is
only sinusoidal function term. Thus, the constant value and thevariation are respectively compared. The constant value, whichmeans amplitude, and the variation, which means phase, arerespectively compared in Eqs. (11–12)(11)

Vout I out  CVm 2(12).

From Eq. 11, the initial phase difference of the neutral point
current  is calculated. Moreover, from Eq. 12, the amplitude of
the capacitor voltage Vm is determined. Then, Vm is represented
as

Vm 

Vout I out
C

(13).

Note that Vm should not exceed the half of Vdc because theDC-link voltage is the sum of the two capacitor voltages,

0A

10 ms/div

Fig. 4. The relationship of the absolute current reference value between the
output current and neutral point current. The neutral point current control
and the output current control interrupt each other in CCM during theshaded area in Fig. 4.
Command value
Average value

inReal current

By Eq. 9 and Eq. 13, the neutral point current which flows to
the neutral point is calculated asin  2 Vout I out C cos  t  
4


i n*

|iout*| , |in*|[A] cos(2t )  sin(2t  2 )iout*

4 A/divt
Zero-current interval(14).

By controlling the neutral point current, the capacitor
voltages are oscillated according to Eq. 5–6 to absorb the power
ripple.
Figure 4 shows the absolute current reference relationshipbetween the output current and the neutral point current. Whenthe single-phase T-type NPC inverter employs CCM as in theconventional method [15], the neutral point current control andthe output current control interrupt with each other during theperiod where the command value relationship is |in*|>|iout*| asshown in the shaded period in Fig. 4. When the neutral point
current flows during the period, the current which is larger thanthe output current reference value, flows to the grid-tied inductor.
Thus, the output current is affected by the neutral point current.
In the conventional method, only the output current is controlled
in the period in order to avoid the interference. Consequently,
the power ripple still exists in the DC-link since the neutral point
current is different from the reference value.
Therefore, DCM is applied for the current controls of thesingle-phase T-type NPC inverter drawn in Fig. 2 in order toavoid the interference. In DCM, there are zero-current intervals,
where the inductor current becomes zero and all switching
devices are turned off. For this reason, the interference of thetwo current control is eliminated by flowing each current during
the zero-current interval of the other current. In particular, theneutral point current flows during the zero-current interval of thegrid current and vice versa. If the zero-current interval of eachcurrent is designed to be long enough to flow the other current,
two currents can be independently controlled simultaneously.
Figures 5 and 6 show the current waveform of the neutral
point current and the output current of the T-type NPC inverterdrawn in Fig. 2 with DCM. As shown in Fig. 5, the zero-current
interval is produced after flowing the neutral point current in.

One switching period

Fig. 5. Waveform of in operated in the DCM
Command value
Average valueiLtiout control

Real current
One switching period

Fig. 6. Waveform of iL operated in the DCM

By the control of this neutral point current according to Eq.
14, the two capacitor voltages oscillate according to Eq. 5–6,
achieving the power decoupling control. As shown in Fig. 6, theinductor current iL flows during the zero-current interval of theneutral point current so that the sum of iL and in results in thesinusoidal output current. Thus, two currents are simultaneously
controlled using only one of connected inductors in series,
leading to the achievement of the power decoupling control andthe sinusoidal output current control simultaneously.
IV. SWITCHING PETTERN SELECTION FOR INDUCTOR CURRENT
REDUCTION
In the proposed method, there are two switching patterns toachieve the operation. In this chapter, the features of eachswitching pattern, and the selection method of the suitableswitching pattern, are explained. Note that the current reference
values are assumed to be “iout* = +, in* = –” in this chapter.

Fig. 7 and 8 show the example switching patterns. The
output current is controlled by the switching pattern in Fig. 7.
Thw neutral point current is controlled by the switching patternsin Fig. 8. Note that the neutral point current flows to the
inductor i.e. output side. The neutral point current interrupts theoutput current control. Therefore, interruption is compensatedby the output current control. Fig. 9 shows the general form of
the inductor current. At first, the output current is controlled byusing the pattern which shown in Fig. 7. After that, the neutral
point current is controlled using the zero-current period of iout
control. However, as shown in Fig.8, there are two of switchingpatterns to achieve the operation. Output current and neutral
point current does not change even which switching pattern isselected. However, the peak value and the RMS value of theinductor current are different. Hence, the inductor current issuppressed by selecting the optimal switching pattern. In thispaper, the two of current control method are called “1st method”
and “2nd method”.
“1st method”: the switching pattern that the polarity of capacitor
voltage and output voltage becomes same, is selected.
“2nd method”: the switching pattern that the polarity of theneutral point through inductor, becomes same as the polarity ofthe output current, is selected.

current. Thus, higher compensation value of the output current
is required in order to eliminate the interruption due to theneutral point current control.
Consequently, the inductor current is suppressed as theminimum value by applying 2nd method during the period that
2nd method is applicable.
V. SIMULATION RESULT
Table I shows the simulation parameters. Fig. 10 shows the
harmonic analysis of the input current, whereas Fig. 11(a) and(b) show the waveforms of the input current, the capacitor
voltages, the output voltage, the inductor current without andwith the proposed power decoupling control respectively.
As shown in Fig. 11(a), the input current oscillates at twice
the grid frequency because the use of the small capacitance of C
leads to the high current ripple on DC-link according to Eq. 3.
On the other hand, as shown in Fig. 11(b), the current ripple ofthe input current is reduced significantly compared to that in Fig.
11(a) because the capacitor voltages VC1 and VC2 oscillate toabsorb the power ripple in the proposed power decoupling
method. In addition, by the proposed method, the component of
100 Hz, i.e., twice the grid frequency is reduced by 97.1%

As the feature of the 2nd method, the inductor current isreduced compared to the 1st method. The compensation value ofthe output current becomes minimum because the polarity of theneutral point current is same as the output current As shown in
Fig. 9. However, the 2nd method cannot be used when thecapacitor voltage is lower than the output voltage because thepolarity of capacitor voltage and the output voltage is opposite.

+

C1

S5,6

S2

S1

vac

L2
+

C2

S4

S7,8

S3

Fig. 7. The switching pattern to control the output current as positive
polarity.

iL

in

Vdc

iL
L1

in

Vdc

On the contrary, the 1st method achieves the current control
at all time because the polarity of the capacitor voltage andoutput voltage are same. However, the polarity of neutral point
current through inductor is opposite to the polarity of the output
+

S1

S2

S5,6

C1

+

vac

S2

S5,6

C1

S1

iL

in

Vdc

vac
+

C2

S7,8

S4

+

S3

C2

S4

S7,8

S3

(a). inductor current increases(b).inductor current decreases
Fig. 8 The switching pattern to control the neutral point current as negative polarity. The polarity of neutral point current becomes same even which pattern isselected. However, the polarity of inductor current is different.

iL

iL
Outputcurrent

No compensation value
Output current

Compensation

iout control

in control
iout control

in control(a). small inductor current due to 2nd method(b).large inductor current due to 1st method
Fig. 9. The inductor current waveforms. The output current and neutral point current become same, whereas the RMS and peak value of the inductor current
is different.

compared to that result without the power decoupling method
whereas the THD of the output current is 1.31%.

Figure 13 shows the harmonic analysis of the input current,
whereas Figs. 14 (a) and (b) show the waveforms of the output
voltage, the output current, inductor current, and the input
current with and without the proposed power decoupling method
at a 500 W load respectively. From Fig. 14 (b), the form of theinductor current is same as the ideal inductor current as shownin Fig. 12(b). In addition, the output voltage is controlled in thesinusoidal waveform at 50 Hz whereas the input current iscompensated by the proposed method. The THD of the output
current is 1.9%. Furthermore, in Fig. 13, by using the proposed
method, the component of 100 Hz, i.e., twice the grid frequency
is reduced by 67.7% compared to that result without the power
decoupling control.

Figure 12 shows the inductor current waveforms when the
output power is 500 W. As the current control method, only the
1st method is employed in Fig. 12(a), whereas both the 1st andthe 2nd method employed in Fig. 12 (b). The RMS value of iL
shown in Fig. 12 (b) is 19.0 A whereas Fig. 12 (b) is 38.8 A
regardless the common output current and the neutral point
current. Thus, the RMS inductor current of Fig. 12 (b) is reduced
by 50.0% compared to the result of Fig. 12 (a) whereas theoperation capability does not change.
VI. EXPERIMENTAL RESULTTABLE I SIMULATION PARAMETERS
Parameter
Output power
Output voltage
Output frequency
DC link voltage
Capacitor
Inductor

Symbol
Pout
vac
fout
Vdc
C1,C2
L1,L2

Value
500 W
100 V
50 Hz
400 V
120 F22.5 H

Inductor Current [%]100%:100Hz component
without Power Decoupling

Table II shows the experimental parameters. The
experiments have been done with an open-loop. In general,
percent impedance %ZL of the grid-tied inductor is designed
around 5%. However, as shown in Table II, the grid-tied
inductor of the prototype is 45H (series connection of the two
inductors 22.5 H). Hence, the %ZL is 0.16%. Furthermore, theadditional inductor which is required for conventional power
decoupling control is not needed.

VII. CONCLUSION
This paper proposed the power decoupling method based on
DCM for T-type NPC inverter without any additional passive
components and switching devices. The proposed control
method achieved the low ripple current at the DC-link and the
low grid current THD. Thus, the reduction of the system size due
to DCM, and the long lifetime are expected from the active
power decoupling method. As the simulation result, the current
ripple of the input current at twice the grid frequency is reducedby 90.2%. Moreover, the output current THD is 1.87%.

100

With Power Decoupling
Without Power Decoupling
Reduced by 97.1%

10
1

0.1
0 1 2 3 4 5 6 714
8 9 1012
harmonic order

16

18

20

Idc[A]
vout[V]

VC1,VC2[V]0V(100 V/div)

0V(100 V/div)

0A(50 A/div)

iL,in[A]

Idc[A]

200 V(100 V/div)

vout[V]

200 V(100 V/div)

2A(1 A/div)

iL,in[A]2A(1 A/div)

VC1,VC2[V]Fig.10. Harmonic components of input current

0A(50 A/div)

5 ms/div

5 ms/div(a) Operation without power decoupling control.
(b) Operation with power decoupling control.
Fig. 11. Waveforms of input current, capacitor voltages, output voltage and output current.in steady state.

(60 A/div)
0A

TIME:5 ms/div(60 A/div)

TIME:5 ms/div

0A(a)only 1st method is applied(b)1st and 2nd method is applied
Fig. 12. Inductor current waveform operated in DCM.

Parameter
Output power
Output voltage
Output frequency
DC link voltage
Capacitor
Inductor

Symbol
Pout
vac
fout
Vdc
C1,C2
L1,L2

Value
500 W
100 V
50 Hz
400 V
166 F22.5 H

Inductor Current [%]100%:100Hz component
without Power Decoupling

TABLE II INVERTER PARAMETERS

100

Reduced by 67.7%

With Power Decoupling
Without Power Decoupling

10
1

0.1
0 1 2 3 4 5 6 714
8 9 1012
harmonic order

16

18

20

Fig.13 Harmonic components of input current

Output Voltage vout (100 V/div)

Output Voltage vout (100 V/div)▶

▶

Output Current iout (10 A/div)

Output Current iout (10 A/div)▶

▶
▶

Inductor CurrentiL (50 A/div)

Input Current
Iin (2 A/div)

Inductor CurrentiL (50 A/div)▶
Input Current Iin (2 A/div)▶

▶

(a) Operation without power decoupling control.
(b) Operation with power decoupling control.
Fig. 14. Waveforms of input current, output voltage, output current and inductor current in steady state.

In the proposed method, there are two switching patterns
to achieve the active power decoupling control. However, theinductor current is different between the two switching patterns.
Thus, the RMS value of the inductor is suppressed by selecting
the suitable switching patterns.
As the experimental result, the output current is controlled as
50 Hz sinusoidal waveform whereas the current ripple of the
input current is suppressed compared to that without power
decoupling control. Furthermore, the form of the inductor
current is same as the ideal current as shown in Fig. 12(b) tosuppress the inductor current. As a result, the second harmonics
component of the input current is reduced by 67.7% comparedto that result without the power decoupling control whereas theoutput current THD is 1.9%.
In the future work, operation at a rated load of 1 kW with acritical current mode, and loss analysis with both simulation andexperiment will be considered in order to clarify the accurate
efficiency.
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