Isolation System with Wireless Power Transfer
for Multiple Gate Driver Supplies
of a Medium Voltage Inverter
Keisuke Kusaka, Koji Orikawa and Jun-ichi Itoh

Kazunori Morita and Kuniaki Hirao

Dept. of Energy and Environmental
Nagaoka University of Technology
Nagaoka, Niigata, Japan
kusaka@stn.nagaokaut.ac.jp

Research & Development Group
Meidensha Corporation
Numazu, Japan

Abstract— In this paper, a multiple wireless power transfer
system for multiple gate driver supplies of a medium voltage
inverter is developed. The proposed isolation system achieves agalvanic isolation with an air-gap of 50 mm using a wirelesspower transfer with magnetic resonance coupling. It easily
respects the standard of galvanic isolation, which is establishedby International electrotechnical commission (IEC). Moreover,the power is supplied from one transmitting board to six gate
drivers without a solid magnetic core. In this paper, the isolation
system is developed and tested. It is clarified that the isolation
system transmits power of not less than 300 mW to each gate
drivers beyond an air-gap. However, sum of the output power ofthe each receiving board are limited up to approximately 3.5 W
because of a voltage drop in the equivalent series resistances of
the transmission coils.

isolation system, a single-chip DC-isolated gate drive IC has
been demonstrated [3-5]. It supplies power using a microwave
from a bottom layer of a sapphire substrate to a top layer. In
this method, galvanic isolation is achieved by the sapphire
substrate. It can downsize an isolation system vastly. However,it does not satisfy the safety standards because both of an
isolation distance and a creepage distance are not enough.
Meanwhile, J. W. Kolar et al. proposed the isolation systemwith using a printed circuit board (PCB) [6]. It achieves agalvanic isolation with a coreless transformer. In this system,
one transmitting side transmits power to a receiving side oneby-one (1×1). Thus, the systems are also required at each gate
driver supply. Therefore the reduction of the volume is limited.
In this paper, a galvanic isolation system with a multiple
wireless power transfer system with magnetic resonancecoupling for a medium voltage inverter is developed. The
isolation system transmits power from one transmitting board
to six receiving boards (1×6) beyond an air-gap of 50 mm.
Because all of the isolation system is constructed by the PCBs.
The isolation system contributes a cost reduction and adownsizing of the isolation system. Besides, the insulation
with an air-gap of 50 mm easily respects IEC standard of aclearance and a creepage distance, when the system voltage ofthe inverter is 6.6 kV. Moreover, the air-gap of 50 mm reduces
a common-mode current, which is induced by high dv/dt of themedium voltage inverter, to a small-signal circuit.

Keywords— Galvanic isolation, Medium voltage inverter,
Wireless power transfer, Magnetic Resonance CouplingI.

INTRODUCTION

In recent years, system voltage of a three-phase medium
voltage inverter for a motor drive system is rising to 3.3 kV
and 6.6 kV. In the medium voltage inverter, galvanic
isolations are required at gate driver supplies. The safety
standards, which are established by IEC [1], have to besatisfied for safety. The safety standards require a minimum
clearance of 14 mm and a creepage distance of 81 mm when asystem voltage of the inverter is 6.6 kV, a comparative
tracking index (CTI) is 100 ≤ CTI < 400 and a pollution
degree is two [1].
In general, isolation transformers with solid magnetic cores
are used for a galvanic isolation. However, it causes anincrease in a cost because it is typically custom-built.
Moreover, the isolation transformer is huge in order to obtaina high isolation voltage. For example, a typical dimension of
the isolation transformer, which have an isolation voltage of
20 kVrms for 10 s, are 200 mm × 200 mm × 200 mm at aweight of approximately 5.5 kg [2]. These transformers areplaced at each gate driver supplies.
In order to achieve a cost reduction and a downsizing of the

II.

PROPOSED ISOLATION SYSTEM

A. System Configuration
Fig. 1 shows the system configuration of the developed
galvanic isolation system for a medium voltage inverter. The
isolation system consists of the transmitting board and six
receiving boards. The power consumption in the gate drive
units (GDUs) is supplied through the isolation system from thelow-voltage power supply of 48 V in a medium voltage
inverter.

B. Rated Power of Gate Driver Supplies
The isolation system transmits the power consumption of
the gate drivers. In this subsection, required power of the eachgate driver is calculated.
Fig. 4 shows the five-level diode-clamped multilevel
inverter as a medium voltage inverter, which has a rated
output voltage of 6.6 kV and a rated output power of 1 MVA[13]. Each switching device is a string of three 1.7-kV IGBTs
connected in series. The power consumption of a gate
resistance PG of an IGBT is calculated by (1) where fc is aswitching frequency, Qg is total gate charge and VGE are gateemitter voltage of the IGBT.

()

PG = f c + Qg + − Qg ( + VGE + − VGE )(1)

From eq. (1), the power consumption is calculated as about

50 mm

m
m

150 mm

15
0

m
m

15

0m

m

Fig. 1. System configuration of the developed isolation system. Poweris transmitted from the transmitting board to the six receiving boards
by a wireless power transfer with magnetic resonance coupling.

60

Incidentally, magnetic resonance coupling achieves awireless power transfer with a resonance with a high-quality
factor Q of the transmission coils in a middle-range
transmission distance [7-12]. In this system, 2.18 MHz is usedas the transmission frequency because a high-frequency
transmission is required in order to downsize the transmission
coils. Then the isolation system does not require a transformer
with a magnetic core. In the conventional system, thetransformer with a magnetic core prevents an isolation systemfrom a cost reduction and a downsizing. In contrast, theisolation system is constructed only by the PCBs in thissystem. The PCBs can be manufactured easily in comparison
with the transformer.
Fig. 2 shows the positional relationship of the transmittingboard and the receiving boards. The maximum size of thesystem is constrained up to 300 mm × 150 mm × 150 mm fora reason of the space limitations of the medium voltage
inverter. The receiving boards are placed at top and bottom of
the transmitting board. Each distance between the receiving
boards and the transmitting board is kept at not less than 50
mm for the galvanic isolation. It contributes the high isolation
voltage and a low common-mode current through leakage
capacitances. It is enough to fulfill the safety standards of IEC[1] when the operating voltage of the medium voltage inverteris 6.6 kV.
The transmitting boards consist of a high-frequency
inverter, a series resonance capacitor and a transmitting coilfor a wireless power transfer system. The inverter is operated
by a square-wave operation with an output frequency of 2.18
MHz because high-frequency operation is necessary in orderto downsize the transmission coils on the PCB. On the other
hand, the receiving boards consist of a receiving coil, series
resonance capacitor and a diode bridge rectifier.
Fig. 3 shows the photograph of the prototype of theisolation system. Each board is placed according to the Fig. 2.
Table I indicates the specifications of transmitting coil andreceiving coils.

Fig. 2. Placements of each boards of the proposed isolation system.
The receiving boards are placed up and down to the transmittingboard. Each board are placed keeping an air-gap of 50 mm.

Fig. 3. Photograph of proposed isolation system.

60 mW where a switching frequency of the medium voltage
inverter is 1 kHz, total gate charges ±Qg are ±1000 nC andgate-emitter voltage is ±15 V. Note that the values, which isused in this calculation, are typical value of an IGBT (VCE =1700V, IC = 150 A). Considering a power loss in a gate drivercircuit, power of at least 120 mW is required per one receiving
board as the output power of the isolation system.

III.

FREQUENCY CHARACTERISTICS WITH
ELECTROMAGNETIC ANALYSIS

In this section, the part of the wireless power transfer in theisolation system is analyzed with the electromagnetic analysis
with Agilent advanced design system (ADS). In ADS, a 3-D
model is analyzed by the momentum method.
Table I shows the specifications of the coils for the analysis.
The isolation system transmits power with the wireless powertransfer with the series resonance capacitors. The windings ofthe coils are made up on the surfaces of the PCBs. In thesystem, series resonance capacitor in the transmitting side andseries resonance capacitors in the receiving side, which iscalled as S/S resonance, are used. The resonance capacitors on
the transmitting board and the receiving boards are 130 pF and
70 pF, respectively. The wireless power transfer with S/Sresonance type has an advantage compared to other method
such as a series resonance and parallel resonance (S/P),
parallel resonance and parallel resonances (P/P) from thestandpoint of the variation of a coupling coefficient and a load.
The load characteristics do not affect the resonance frequency
in S/S resonance. It is suitable characteristic for the isolation
system because the loads have different value in each GDU in
the isolation system. Also, the coupling coefficient does notaffect the resonance frequency in the S/S resonance. The
coupling coefficients vary among the receiving boards. It
means that, the isolation system with S/S resonance can be
operated at a constant operating frequency without a complex
control.
Fig. 5 presents the definition of S-parameters. The Sparameter is one of the methods to express the characteristics
of a multi-terminal circuit. The relationship between a squareroot of input power and a square root of output power isexpressed by the matrix shown as eq. (2) where a1 is a squareroot of travelling power, b1 is a square root of reflected powerin the primary side, a2 is a square root of travelling power andb2 is a square root of reflected power in the secondary side. In
this paper, the S-parameters are used to analyze thecharacteristics of the isolation system.

⎛ a1 ⎞ ⎛ S (1,1) S (1, 2) ⎞⎛ a2 ⎞⎜⎜ ⎟⎟ = ⎜⎜⎟⎟⎜⎜ ⎟⎟⎝ b1 ⎠ ⎝ S (2,1) S (2, 2 )⎠⎝ b2 ⎠(2)

Fig. 6 shows the definition of S-parameters in the system.
In this consideration, a S-parameter S(n,0) is especially
focused. The S-parameter S(n,0) is the ratio of a square root of
power from the transmitting board to the receiving boards,
where n is the number of the receiving boards (n = 1, 2, ···, 6).
In the isolation system, the power flow is limited to one-way;from the transmitting board to the receiving board. Thus, thetransmission from the receiving boards (#n) to the transmittingboard (#0) can be ignored.
Fig. 7 shows the S-parameters S(n,0) of the system. The Sparameters S(2,0) and S(5,0) reach to -9 dB at the resonancefrequency. It means that the each ratio of the output power ofFig. 4. Assumed 6.6 kV, 1 MVA five-level diode-clamped medium
voltage invertere.
Table I. Specification of coils.
(a) Transmitting coil.
Items
ValuesNumber of turn

12

turn

Width

250

mm

Depth

40

mm

Line width

1mm

Gap between windings

0.7

mm

Thickness of PCB

1.6

mm

Film thickness of copper

70

μm

(b)
Items

Remarks
Short type

FR-4

Receiving coils.
ValuesNumber of turns

40

turn

Outer diameter

44mm

Inner diameter

22mm

Gap between windings

0.4

mm

Line width

0.2

mm

Thickness of PCB

1.6

mm

Remarks
Short type

FR-4

⎛ S (i, i ) S (i, j ) ⎞⎜⎜⎟⎟⎝ S ( j , i ) S ( j , j )⎠Fig. 5. Definition of S-parameters.

the receiving boards (#2 and #5) to input power of thetransmitting board are 13.2%. In contrast, the S-parameters
S(1,0), S(3,0), S(4,0) and S(6,0) cannot reach to -10 dB. Thus,
the each ratio of the output power of the receiving boards (#1,
#3, #4 and #6) to input power calculated as 7.4 %. The nonuniform transmitted power is caused by the difference in the
coupling coefficient among the boards. The receiving boards,
which are placed in center (#2 and #5) are coupled to thetransmitting board (#0) strongly compared to other boards.
Fig. 8 shows the effect between the abutting receiving

0
-20
-40
-60
-80

boards. The transmission between the receiving boards issignificantly small than the transmission between thetransmitting board (#0) and the receiving boards (#1-6). For
example, the transmission from the transmitting board (#0) to
the receiving board (#1) S(1,0) reaches to -11.4 dB. On theother hand, the transmission from the receiving board (#2) tothe receiving board (#1) S(1,2) is a -17.3 dB. Thus, thecoupling between the receiving boards can be ignored in thesystem.
Fig. 9 shows the simulated transmission efficiency. The
simulation is held without the converters such as an inverter
and a rectifier. The transmission efficiency expresses the ratioof sum of the received power on the receiving boards to thetransmitted power from the transmitting board. Thus, thetransmission efficiency ηΤ is calculated by (3). It should be
noted that the output impedance of the power supply and theload impedance, which are used for an analysis, are 50 Ω.

ηT =∑ S (n, 0)

2(3)

n =1, 2 ,"6IV.

TIME-DOMAIN ANALYSIS

A. Equivalent Circuit
Fig. 10 presents the equivalent circuit of the wireless powertransfer [14-15] where r0-6 are the equivalent series resistancesof the windings and C0-6 are the series resonance capacitors.
The equivalent circuit of the multiple wireless power transfer

2

2.5

3

0
-20
-40
-60
-80

-100

1

1.5

2

2.5

3

Fig. 8. Effect between the abutting receiving board. The transmission
between the receiving boards such as S(1,2) is smaller than thetransmission from #0 to #1.

60
50
40
30
20
10
0

At a resonance frequency of 2.18 MHz, the totaltransmission efficiency increases drastically because magnetic
resonance coupling achieves the efficient wireless powertransfer with a high quality factor Q. The maximum
transmission efficiency reaches to 53.6% at 2.18 MHz.
In the isolation system, low transmission efficiency is
accepted because the power loss in the proposed isolation
system is extremely low compared to the power loss of a
medium voltage inverter, typically. Thus, the power loss of theisolation system can be ignored.

1.5

Fig. 7. Positional dependence of the receiving boards on the Sparameters. The S-parameters between the transmitting board #0 andthe receiving board #2, 5 are larger than the S-parameter between thereceiving board #0 and #1, 3, 4, 6.

S(1,0), S(1,2) [dB]Fig. 6. S-parameters in the isolation system. All of the boards arenumbered from #0 to #6. All of the analyses with S-parameters in thispaper are held with a characteristic impedance of 50 Ω.

1

1

1.5

2

2.5

3

Fig. 9. Transmission efficiency of the isolation system. The efficiency
is analyzed from the 3-D model with ADS.

is obtained as transformers with multiple windings where k is
the coupling coefficient among the each winding, which isexpressed as (4). Note that, the subscript indicates the numberof the transmitting board.

⎛ 0
⎜
⎜ k10
k = ⎜ k 20⎜
⎜ #⎜⎝ k 60

k 01
0
k 21
#
k 61

k 02 " k 06 ⎞⎟k12 " k16 ⎟0 " k 26 ⎟⎟
# % # ⎟⎟
k 62 " 0 ⎠(4)

If the self-inductance of the each winding L is expressed as(5), the leakage inductance Lle and magnetizing inductance Lmare provided as (6) and (7), respectively. It should be noted
that, kij is equals to kji (i, j = 0, 1, ···, 6).
L = (L0

L1

L2

L3

L4

L5

L6 )

(5)

L le = L − L0 (1 1 1 1 1 1 1)k(6)

L m = Lk

(7)

The parameters; the self-inductance, the series resonancecapacitors and the coupling coefficients are introduced by the
analysis results with ADS at the resonance frequency of 2.18
MHz. Table II shows the derived parameters. In Fig. 10, thecoupling coefficients among the receiving boards are ignored
because the effect of these coupling is significantly small. It
means that the receiving boards are only magnetically coupled
to the transmitting board.
Fig. 11 shows the comparison results of the frequency
characteristics of the F-parameters. The F-parameters arecompared between the equivalent circuit and the 3-D modelwith ADS in frequency characteristics. ADS indicates thecircuit characteristic as S-parameters, typically. Thus, theanalysis results are converted to the F-parameters according toeq. (8).
⎛Vi ⎞ ⎛ A B ⎞⎛V j ⎞
⎜ ⎟=⎜
⎟⎜ ⎟
⎜ I ⎟ ⎜ C D ⎟⎜ I ⎟⎠⎝ j ⎠⎝ i⎠ ⎝=()⎛ (1 + S ii ) 1 − S jj + S ji S ij
1 ⎜⎜1
2S ji ⎜ (1 − S ii ) 1 − S jj − S ji S ij
Z i
⎝

{

(

)

}

{(1 + S )(1 + S ) − S S }Z ⎞⎟
{(1 − S )(1 + S ) + S S } Z ⎟⎟
ii

jj

ji

ij

ii

jj

ji

ij

(8)

j

Zi ⎠

B. Operation modes
Fig. 12 shows the equivalent circuit with converters for the
analysis of the operation modes of the system. The series
resonance capacitors are designed according to eq. (9) whereωsw is 2πfsw. The resonance capacitance C0 on the transmittingboard is selected in order to obtain a resonance with the selfinductance of the transmitting coil. Similarly, the resonancecapacitances on the receiving boards are selected to resonate
with the each self-inductance L of the receiving boards.

C=

2
ω sw
L

120

3-D model (ADS)
Equivalent circuit (PLECS)

B10

80
D10

40

A10

0
C10

-40
-80

1

2

3

4

Frequency f [MHz]Fig. 11. Comparison results of frequency characteristics between the
3-D model with ADS and the equivalent circuit mode.

j

The equivalent circuit model shows the good agreementwith the 3-D model. The error is caused by the unconsideredparameters; parasitic capacitances between the layers of the
PCB, parasitic capacitances between the windings and adielectric loss of the PCB. These parameters cannot be derived
even by the simulation. Furthermore, the error occur owing to
the non-linearly characteristics of an impedance of thetransmission coils because the impedance of the coils are notproportional to a frequency in the high-frequency region.
However, the difference does not cause a fatal error on thetime-domain analysis. Thus, the operation model is discussed
using the equivalent circuit model in the next subsection.

1

Fig. 10. Simplified equivalent circuit of the proposed isolation
system.

(9)

Table II. Derived parameter value from the 3-D analysis.
Items
SymbolValuesEquivalent series
resistances (ESRs)

Self-inductances

Board #0

r0

12.9

Ω

Board #1

r1

11.7

Ω

Board #2

r2

11.6

Ω

Board #3

r3

11.7

Ω

Board #4

r4

11.7

Ω

Board #5

r5

11.6

Ω

Board #6

r6

11.7

Ω

Board #0

L0

41.8

μH

Board #1

L1

76.5

μH

Board #2

L2

75.9

μH

Board #3

L3

76.5

μH

Board #4

L4

76.5

μH

Board #5

L5

75.9

μH

Board #6

L6

76.5

μH

k01

0.016

k02

0.026

k03

0.025

k04

0.016

k05

0.026

k06

0.025

Coupling coefficients

A. Fundamental Verifications
Fig. 15 shows the operation waveforms with the prototype,which is shown in Fig. 3. Fig. 15 (a) is the waveforms withno-load condition. Fig. 15 (b) is the loaded waveforms with aresistance load of 107 Ω where the input power of theisolation system is 31.3 W. In this experiment, resistances areused instead of the gate drivers for simplicity.
From the experimental waveforms, it is demonstrated thatthe wireless power transfer with an air-gap of 50 mm isachieved. An air-gap of 50 mm behaves as an isolation
distance. Therefore, it is clear that the isolation system
satisfies the safety standards, which is established by IEC[1], when a system voltage of the medium voltage inverter is
6.6 kV.
When the load is not connected, the rectifier input voltage
vin1 is a sinusoidal because the diodes in the receiving boards
do not turn-on. In contrast, the rectifier input voltage becomes···

EXPERIMENTAL RESULTS

···

V.

···

Fig. 13 shows the vector diagrams of the isolation system.
Owing to the resonance based on eq. (9), the inverter voltage
and the current are in phase. The terminal voltage of theresonance capacitor and the inductance, become extremely
high with opposite directions. Focusing on the receiving side,
the resonance capacitor C1 and the leakage inductance Lle1
resonate.
Fig. 14 illustrates the simplified operation waveforms of
the isolation system.
1) Mode I
The inverter current iinv starts to flow through the
MOSFETs and the coils because the MOSFETs S1 and S4 areturn-on. Owing to the resonance, the input current becomes asinusoidal and it is in phase to the inverter output voltage vinv
during a period.
2) Mode II
The inverter current iinv commutates to the diodes D2 andD4 because the all of MOSFETs turn-off during the dead-time
Td. If the dead-time is significantly short compared to theswitching period, the MOSFETs achieve a zero current turnoff owing to the current resonance. After the half of theswitching period, the inverter current crosses zero andcommutate to the diodes D1 and D3. The inverter output
voltage is decided by the polarity of the inverter output current.
3) Mode III
The MOSFETs S2 and S4 turn-on. The inverter current
commutates to the MOSFETs S2 and S4 from the diodes. Also,
the inverter current is sinusoidal owing to the resonances.
4) Mode IV
The MOSFETs S2 and S4 have turned-off at the start of thismode. Owing to the current resonance, the MOSFETs turn-off
around a zero current. It contributes a reduction of theswitching loss. The inverter currents commutate to the diodes
D1 and D4. The directions of the inverter current change to a
positive from a negative.

Fig. 12. Equivalent circuit with converters. The ideal transformers areomitted for simplicity.

Fig. 13. Vector diagram of the wireless power transfer. The series
resonance capacitor C0 resonates with the sum of leakage inductanceLle0 and the magnetizing inductance Lm. The self-inductance L2 and theseries resonance capacitor C1 resonate.
Td
Td
Gate onS1,4 offon
S2,3 off
vinv

0

iinv

0

is1

0

is2

0

iD1

0

iD2

0

vin1

0

iin1

0

Mode I
T
2

II

III

IV

T
Fig. 14. Simplified operation waveforms. The MOSFETs achieve azero-current turn-off because of the current resonance.

a trapezoidal wave when the load is connected. This is causedby the sinusoidal-rectifier input current iin1.
As an output voltage, a DC voltage VDC1 is obtained despite
a load value. However, the output voltage decreases when theload is connected. It is caused by the voltage drop on theequivalent series resistances of the transmission coils r0-6.
Incidentally, the others receiving boards simultaneously output
the DC voltage. In this paper, the waveforms are omitted dueto the page limitations.
Fig. 16 shows the output power of each receiving board.
The output power varies among the position of the receiving
boards. The difference between the output power is caused by
the accuracy of the chassis and differences of the coupling
coefficients. In the experimental setup, the output power of thereceiving boards (#1, 2, 3) are smaller than that of (#4, 5, 6),
respectively because the difference in the transmission
distance occur between the upper boards and lower boards.
Furthermore, the output power of the receiving boards #2 and#5 are larger than that of adjacent boards, respectively. Thus,
the output power of #5 has maximum output power. The
minimum output power reaches to 320 mW. It is confirmedthat the isolation system supplies the power, which is requiredin the gate driver supplies.
Fig. 17 shows the total output power Pout_total characteristics.
The total output power is calculated as (10).
Pout _ total =∑ P (n )out
n =1, 2 , ", 6(a)No-load.

(10)

The output power increases as a load resistance decreases
in the interval of the load resistance from 125 Ω to 750 Ω.
However, the output power stops to increase when the small
resistance such as 107 Ω is connected because the voltage
drop on the equivalent series resistances of the transmission
coils increases owing to the increment of load current.
B. Operation with Gate Drivers
In this subsection, the prototype with the gate drivers is
tested. The gate drivers, which are operated at a switchingfrequency of 1 kHz, a gate-emitter voltage of ±15V, areconnected as a load. Note that DC/DC converters as voltage
regulators are connected at the front end of the gate drivers.
The DC/DC converter output voltages of ±15 V. Moreover,the photocouplers (Toshiba, TLP250) are used in order todrive the IGBTs in spite of the deficient isolation distancebecause an isolation of the PWM signal is not a main topic of
this paper.
Fig. 18 shows the operation waveforms where thecapacitors of 33 nF is connected instead of the IGBTs. Even if
the gate drivers are connected as a load, the operation systemachieves the wireless power transfer beyond an air-gap of 50
mm.
From the experimental results, it is confirmed that the
proposed isolation system can be used as an isolation system
for the medium voltage inverter.

(b) Loaded.
Fig. 15. Operation waveforms. The load resistance of 107 Ω isconnected as a load instead of a gate driver supplies.

1200
1000
800
600
400
200
0
Fig. 16. Operation waveforms. The output power varies among theposition of receiving boards. The difference of output power is causedby the difference in the accuracy of the chassis.

VI.

CONCLUSIONSIn this paper, the isolation system for gate drivers of a
medium voltage inverter with the system voltage of 6.6 kV isreported. The isolation system transmits the power to the six
gate drivers beyond the air-gap of 50 mm. The isolation

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n=1, 2, ···, 6

REFERENCES

4
Total output power∑ Pout (n) [W]distance satisfies the safety standards in IEC [1]. The isolation
system consists of the only seven PCBs. It contributes a cost
and weight reduction because it does not need a magnetic core.
The experimental results demonstrate that the isolation system
supplies the power of not less than 320 mW to the each
receiving boards. It is enough power to drive the IGBT in themedium voltage inverter.

2
1
0

200

400

600

800

Fig. 17. Output power vs. load resistance. Owing to the voltage dropof the equivalent series resistance of the coils, the total output power
is constrained at approximately 3.5 W.

(a) Gate-emitter voltage with proposed isolation system.

(b) Extended operation waveforms of (a).
Fig. 18. Operation waveforms of the proposed isolation system withthe gate drivers.