Investigation of a Two-stage Boost Converter
using the Neutral Point of a Motor
Jun-ichi Itoh*, Daisuke Ikarashi** Nagaoka University of Technology, 1603-1 Kamitomioka-cho Nagaoka City Niigata 940-2188, Japan

Abstract--This paper discusses about a six-step operation
strategy for a proposed two-stage boost converter using theneutral point of a motor. The proposed converter consists ofa small boost chopper and a three-phase inverter, which has
a boost up function from the use of the leakage inductanceof a motor instead of a boost-up reactor, for a DC to AC
conversion. When the inverter part outputs a square
waveform of 180 degree, the input current has distortion
because the neutral point voltage of the motor is fluctuatedat three times of the output frequency. The input current
distortion because of the voltage fluctuation is reduced to
1/8 smaller by applying a feed forward control. Additionally,in this proposed circuit, the DC current is imposed into thephase current of the motor. Therefore, the influence of theimposed DC current is investigated in a FEM by simulating
a 1.5-kW IPM motor.
Index Terms--Boost converter, Feed forward control,
Leakage inductance, Six-step operation

I. INTRODUCTION
Recently, motor drive technologies that drive with abattery are required in a lot of applications such as
electric vehicles and rail trains [1]. In order to obtain highefficiency, high terminal voltage at the motor is required.
However, a higher voltage will need to increase thevolume of the battery bank and result the cost will
increase. Therefore, the battery is usually connected to aboost converter before an inverter stage.
Fig. 1 shows the conventional DC/AC power converter,
which is composed of batteries, a boost converter and athree-phase inverter. Battery voltage is increased by the
boost converter when the inverter outputs high voltage tothe motor. However, the boost converter requires a large
boost-up reactor which is high cost and bulky. In addition,higher switching frequency is applied in order to reduce
the size volume of the boost-up reactor, however thisstrategy increases the switching losses. As one of solution
to reduce the switching losses under high frequency,
resonance converters are used [2-4]; however the numberof components is increased in the boost-up converter.
The authors have been proposed a reactor free boostup converter that uses the leakage inductance of a motor
instead of a boost-up reactor [5-6]. The proposed
converters can increase the inverter output voltage
without increasing the number of components. However,when the proposed converter is applied to the DC to AC
conversion, the proposed converter is found with thefollowing three problems;(i) The limitation of the inverter output voltage
When the battery is connected to the neutral point ofmotor, the maximum phase voltage of the inverter islimited by the battery voltage.
(ii) Ripple current caused by the Six-step operation
One of the control methods for motor drive is the sixstep operation. The six-step operation of the inverter canincrease the voltage utilization and achieve higher
efficiency than a PWM operation since the switchingfrequency of a six-step operation is lower than the PWM
operation. In the reactor free boost-up converter, when
the six-step operation is applied to the inverter control,
the neutral point voltage of the motor will fluctuate at afrequency which is three times of the output frequency.
Since the battery is connected to the neutral point of the
motor, the input current will contain large ripple due tothe fluctuation of the neutral point voltage of the motor.
(iii) Increasing of the motor loss
In the reactor free boost-up converter, the DC current
is imposed into the phase current of the motor. As a result,the copper loss of the motor is increased. Meanwhile, in
an ideal condition, the zero-phase current in the motor
does not generate motor torque and iron loss because thezero-phase flux is denied with each other. However, themagnetic flux density might be increased by the zerophase current. Influence of the zero-phase current in the
motor has never clarified before.
This paper proposes a two-stage boost converter using
the neutral point of a motor to solve the problems (i) and(ii). The proposed circuit consists of batteries, a threephase inverter which connects to the neutral point of a
motor and a small boost-up converter. The proposed
circuit can reduce the terminal voltage of a boost-up
reactor. As a result, the proposed circuit can reduce thesize of the boost-up reactor and switching losses in the
boost converter. In addition, the current fluctuation at theneutral point of the motor that resulted from the six-step
operation will be compensated by a current regulator anda feed forward control in a small stage boost converter.
The validity of the proposed converter and its control
strategy are confirmed with a simulation and theexperimental results. Furthermore, in consider for theiron loss and the torque ripple, the influence of (iii) isanalyzed in a 2D Finite Element Method (FEM) by
simulating a 1.5-kW IPM motor.
II. CIRCUIT TOPOLOGY
A. Circuit configuration
Fig. 2 shows the proposed circuit configuration. The
proposed circuit is composed of batteries, a three-phase
inverter and a first stage boost converter that connects tothe neutral point of a motor. The inverter in the proposed
circuit comes with a boost-up function when the motor
leakage inductance is utilized. However, the DC current
is imposed to the phase current of the motor in the
proposed circuit.
The proposed circuit can reduce the volume of theboost-up reactor and switching losses in the first stage
boost converter. When a sinusoidal wave triangle carriercomparison method is used to generate PWM pulses, theneutral point voltage of the motor is the same as theneutral point voltage of the DC link part, i.e. half of theDC link voltage Edc1. Therefore, the voltage rating of theboost-up reactor and switching device in the first stage
boost converter is reduced to half of the DC link voltage.
It should be noted that the capacitor C2 that is connectedto the neutral point of the motor is 100 times smaller thanthe DC link capacitor C1 of the inverter.
B. Positive phase sequence equivalent circuit
Fig. 3 shows the positive-phase sequence equivalentcircuit of the proposed circuit. The proposed circuit canbe divided into a positive and a zero sequence
components. The boost converter does not appear in thepositive-phase sequence equivalent circuit due to theconnection of the neutral point of the motor.
Consequently the positive-phase sequence equivalentcircuit is similar to a conventional three-phase inverter.
During the PWM control, when a sinusoidal modulation
based on the triangle carrier comparison is used togenerate PWM pulses, the fundamental voltage of theoutput is given by
Vout =

3
2 2

Edc ⋅ a .........................................................(1),where Vout is the output line voltage, Edc is the DC link
voltage and a is the modulation index.
In the proposed circuit, a six-step operation is applied
to the three-phase inverter in order to reduce theswitching loss of the inverter in comparison to a PWM
inverter. During the six-step operation, the switchingfrequency agrees with the output frequency. The output
line voltage becomes a 120 degree square waveform, then,the fundamental voltage of the output is given by
Vout =

6π

E dc ...............................................................(2).

C. Zero phase sequence equivalent circuit
Fig. 4 shows the zero-phase sequence equivalentcircuit. The back electromotive force (EMF) does notappear in the zero-phase sequence equivalent circuit but
only the leakage inductance exists. In zero-phasesequence, the inverter legs could consider as a single leg.
Then the output voltage of the single leg is the same as
the neutral point voltage of the motor. A two-stage boost
up operation is achieved by two choppers, which areconstructed by switch S1, S2 and switch S3 and S4.
However, the leakage inductance of the motor decreases
to 1/3 since the leakage inductance of the motor isconnected in parallel in the zero-phase sequenceequivalent circuit.
On the other hand, the neutral point voltage of theFig. 1. Conventional circuit diagram.

Fig. 2. Proposed circuit diagram.
S3

S1

S5

Edc
u
S2

v
S4

Motor

w
S6

Fig. 3. Positive phase sequence equivalent circuit.
Edc1

S1

Leakageinductance

C1

S3

S2
Edc2

C2
S4

L
Iin
EbatteryFig. 4. Zero phase sequence equivalent circuit.

motor is given by
E dc 2 =1
1(vu + v v + v w ) + E dc1 ................................... (3),23where Edc2 is the neutral point voltage of the motor, vu,
vv and vw are the motor phase voltage and Edc1 is the DC
link voltage.
Therefore, the neutral point voltage of the motor canbe controlled by the zero-phase voltage command. In
particular, the average of the neutral point voltage of the
motor for one switching period is zero when thesymmetrical three-phase sinusoidal waveform is used as
modulation signal. Therefore the neutral point voltage ofthe motor equals to double of the DC link voltage.
Furthermore, sum of the maximum inverter output
voltage command and the zero-phase voltage command islimited by the neutral point voltage of the motor. If thebattery connects the neutral point of the motor directly,
the maximum phase voltage of the motor is constrainedby the battery voltage. However in the proposed circuit,the maximum phase voltage is not constrained by thebattery voltage because the neutral point voltage of the
motor is controlled by the boost converter.

v =0
*

f/v
conv

III. CONTROL STRATEGY
A. Control method of output voltage
Fig. 5 shows the block diagram of the proposed circuit.
The control of the motor current in the proposed circuit is
the same as a conventional three-phase inverter because
the positive-phase sequence equivalent circuit is exactly
similar to a conventional three-phase inverter. The
stabilization control is applied to achieve a V/f control forthe permanent magnetic motor [8]. The stabilization
control is composed of a high-pass filter and a feed backfrom δ-axis current. Additionally, the PWM control isslowly changed into a six-step operation via thetrapezoidal pulse modulation [9]. The transition control in
Fig.5 is a proportional compensation to the amplitude of
the output voltage command because the output voltage isnonlinearly increasing at the trapezoidal pulse area.

vu∗

*

v*

Transition
control

↓
uvw

+
-

∗
v

v

+
-

vw∗

+
-

∗
+ ωl 1

iu

iδ
uvw
↓

iv

－
HPF

s

+

K

×

Stabilization
controliw
Edc1* +
－

－

Edc1

+

PI

+
-

S7
S8
Triangular wave+

Iin
Boost chopper control++

Feed forward
control

1/6Edc1

Iin*+

PI

S1
S2
S3
S4
S5
S6

Vconv*

Fig. 5. Control block diagrams.
TABLE ISIMULATION PARAMETERS

B. Battery current compensation control
In the proposed circuit, when the six-step operation isapplied to the inverter control, the input current isdistorting at three times of the output frequency. The
reason is that the neutral point voltage of the motor has
fluctuation of ±1/6Edc1 with three times of the inverter
output frequency in the six-step operation. In order tocompensate the voltage fluctuation, a feed forward
control is applied to a current regulator in the first stage
boost converter. The voltage fluctuation is estimated by
the DC link voltage and the pulse pattern of the inverter.
IV. SIMULATION RESULTS

B. Loss analysis results
Fig. 7 shows a loss analysis comparison between the
proposed circuit and the conventional circuit during a sixstep operation. The loss analysis was implemented by acircuit simulator (PSIM, Powersim Technologies Inc.) and
DLL files (Dynamic Link Library) [10]. It should be
noted that in this loss analysis, the iron loss of a motor
and a boost-up reactor are neglected. The loss simulation
estimates that the proposed circuit can reduce the boost
converter loss by 25% as compared to the conventional
circuit because the proposed circuit can reduce theFig. 6. Simulation results with six-step operation.

100
80
Loss[W]

A. Fundamental Operations
Table 1 shows the simulation parameters, thepermanent magnet motor is 750 W, 160 V and 90 Hz, andthe battery voltage is 70 V.
Fig. 6 shows the simulation results during a six-step
operation. The motor model is expressed by the backEMF, the leakage inductance and the armature resistance.
From Fig.6, it is confirmed that the fluctuation of the
input current is suppressed by the feed forward control in
the first stage boost converter. In additionally, the DC
link voltage Edc1 is controlled to 200 V by a voltage
regulator.

Input voltage:48[V] Output power:750[W] Switching frequency:90[Hz]
Conventional
circuitConventional
circuit

Proposedcircuit60

Efficiency:89[%]

Switching loss

40

Conduction loss

20

Proposed
Conventional circuit
circuit0

Proposedcircuit

Diode loss

Boost chopper
loss

Inverter loss

Efficiency:90.7[%]

Total loss

Fig. 7. Loss analysis results in simulation.

terminal voltage of a boost-up reactor. Furthermore, italso have been confirmed that the inverter switching loss
is almost zero by the six-step operation. However, theinverter conduction loss is increased by the zero-phasecurrent. As a result, the proposed circuit improves theefficiency by 1.7% as compared to the conventional
circuit.

TABLE II
EXPERIMENTAL PARAMETERS

V. EXPERIMENTAL RESULTS
A. Experimental results with PWM control
Table 2 shows the experimental parameters, thepermanent magnet motor is 750 W, 175 V and 90 Hz,
and the battery voltage is 70 V.
Fig. 8 shows the experimental results during a PWM
control. In this experiment, the output torque iscontrolled to 100%. From Fig. 8, it is confirmed that theneutral point voltage of the motor is increased to doubleof the DC link voltage by the inverter side boost-up
function that utilizes the motor leakage inductance.
Furthermore, the motor current has the DC components
because the motor current is added to the zero-phasecurrent. However, the sinusoidal motor current waveformand sinusoidal line voltage of the motor are obtained. It
should be noted that the output line voltage Vuv isobserved by using a low-pass filter of 1.5-kHz cut-off
frequency to observe the low frequency component
distortion.
Fig. 8. Experimental results with PWM control.

B. Experimental results with Six-step operation
Fig.9 shows the experimental result during a six-step
operation. In this experiment, the output torque iscontrolled to 75%. In Fig. 9(a), a fluctuation occurs in the
input current because the neutral point voltage of the
motor has fluctuation. Fig. 9(b) shows that the ripple in
the input current has been suppressed by the proposed
control.
Fig. 10 shows the harmonic analysis results of the
input current. In Fig. 10(a), the input current is confirmedthat is containing the third harmonic components of theoutput frequency. For Fig. 10(b), the feed forward
compensation is applied and this shows that the input
current ripple has been suppressed. In comparison with
Fig. 10(a), the third harmonic component of the output
frequency in the input current is suppressed by 1/8 times
smaller.
C. Torque Impact characteristic
Fig.11 shows the torque impact characteristic. The
output frequency is 90Hz, and the step increase of torqueis 100%. In Fig.11, stability to the load step can be
confirmed. Furthermore, the DC link voltage Edc1
vibration is suppressed by a voltage regulator.
D. Acceleration characteristic
Fig.12 shows the acceleration characteristic fromPWM operation to six-step operation. The PWM control
is gradually changed into the six-step operation that canbe confirmed by the u-v line voltage. The pulse mode
moves from PWM into the six-step mode entirely
without appearing of rush current.
VI. MAGNETIC FIELD AYALYSIS RESULTS
The influence about the imposed DC current isanalyzed in a 2D finite element method (FEM) by(a) Experimental results without feed forward compensation.

0

DC link voltage Edc1[V]

250[V/div]

Line voltage Vuv[V](LPF)

250[V/div]00Motor current Iu[A]0

5[A/div]

Input current Iin[A]

10[A/div]
10[ms/div](b) Experimental results with feed forward compensation.
Fig. 9. Six-step operation experimental results.

simulating a 1.5-kW interior permanent-magnet (IPM)
motor. Table 3 shows the IPM motor parameters. Two
conditions are considered; first, the motor phase current
is assumed as a sinusoidal rated current without the DC
current. Second condition is with the DC current, where
the DC current is -3.14 A per single phase.

Fig. 13 shows the magnetic flux density distribution of
a 1.5-kW IPM motor. Fig. 13(a) shows the magnetic flux
density distribution where the motor contains only thesinusoidal current. Fig. 13(b) shows the magnetic flux
density distribution where the DC current is included.
From the comparison, this shows that the magnetic flux
density distribution is almost the same in both results.
Therefore, this can be confirmed that the increment of theiron losses caused by the zero-phase current is very low.
Fig. 14 shows a comparison of the motor torquewaveform. When the DC current is added to the output
current, the torque ripple is increased by 10% because thetorque ripple is affected by DC current which contains
the third harmonic components of EMF and the spaceharmonics components. It should be noted that the torqueripple that results from the third harmonics component
can be compensated by the feed forward control.
Fig. 15 shows a comparison of the IPM motor loss
analysis. The copper loss is increased by the DC current
component because the copper loss is proportional to thesquare of the motor current. On the other hand, theincrease of the iron loss can be neglected. It should be
noted that the motor current does not consider as carrier
harmonics components by PWM.

(a) Harmonics analysis without feed forward compensation.

VII. CONCLUSION
A novel two-stage DC/AC power converter connectedto the neutral point of the motor has been proposed. The
proposed circuit can reduce the volume of the boost-up
reactor and switching losses in the boost converter. In theloss analysis, the proposed circuit improves the efficiency
by 1.7% as compared to the conventional circuit. The
basic operation of the proposed circuit is confirmed by
the simulation and the experimental results.
A six-step operation is applied to the proposed
converter with a feed forward control to compensate thefluctuation at the motor neutral point voltage. The
experimental results confirmed that the proposed control
method could suppress the input current distortion by 1/8
smaller.
In addition, the influence about the imposed DC
current is analyzed in a 2D FEM by simulating a 1.5-kW
IPM motor. The analysis shows that the torque ripple is
PWM control

DC link voltage Edc1[V]0Motor current Iu[A](b) Harmonics analysis with feed forward compensation.
Fig. 10. Input current harmonics analysis.

Fig. 11. Torque Impact characteristic.

Transition control

Transition control

100[V/div]
5[A/div]

DC link voltage Edc1[V]0Six-step operation

100[V/div]

Motor current Iu[A]

5[A/div]

Line voltage Vuv[V]（LPF）

250[V/div]00Line voltage Vuv[V]（LPF）

250[V/div]0020[ms/div]

20[ms/div](a) Operation waveform from PWM control to Transition control.
(b) Operation waveform Transition control to six-step operation.
Fig. 12. Acceleration characteristics.

2.5T

TABLE III
IPM MOTOR PARAMETERS
Motor Power
1.5kW
Rated Voltage
180V/phase
Rated Current
6.1A
Rated Speed
1800rpm
Number of Poles
6poles36slots
Number of Stator Slots
Stator Outer Diameter
130mm
Stator Inner Diameter
83mm
Winding Configuration
138turn,series per phase
Rotor Outer Diameter
82.2mmRotor Shaft Diameter
30mm

W

W

U

1.25T

U
V

N

V
W
W

0.0T

increased by 10% and the copper loss is increased by
25%. However, the increment of the iron losses causedby the zero-phase current is very low. It should be notedthat the torque ripple that results from the third
harmonics component can be compensated by the feed
forward control. In future work, the motor will be
optimized and reduced the losses in the zero-phasecurrent.

(a) Without DC current.

Flux of DC(-3.14A)

REFERENCES

1.25T

0.0T
(b) With DC current.
Fig. 13. Magnetic flux density distribution.

15
10
5

Without DC (Torque ripple 45.6[%])Torque[Nm]

0
15
10
5

With DC (Torque ripple 58.9[%])0
5Only DCTorque ripple =Max − Min× 100[%]
Average

0
-5
0

0.002

0.006
0.004
Time[s]0.008

Fig. 14. Torque waveforms in simulation.

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2.5T

Fig. 15. IPM motor loss analysis in simulation.

0.01