Experimental Verification and Analysis of AC-DC Converter with an
Input Impedance Matching for Wireless Power Transfer Systems
Keisuke Kusaka, Jun-ichi Itoh
Nagaoka University of Technology
1603-1 Kamitomioka-machi
Nagaoka Niigata, JAPAN
Tel.: +81 / (258) – 47 – 9563.
Fax: +81 / (258) – 47 – 9563.
E-Mail: kusaka@stn.nagaokaut.ac.jp, itoh@vos.nagaokaut.ac.jp
URL: http://itohserver01.nagaokaut.ac.jp/itohlab/en/index.html
This work was supported by Japan Society for the Promotion of Science; Grant-in-Aid for Scientific
Research (B), 2436016.

Keywords« Wireless power transmission », « Power converters for EV », « High frequency power converter »Abstract
This paper discusses the performance of an AC-DC converter which converts power from 13.56MHz AC to DC in a receiving side of wireless power transfer systems. The wireless power transfer
systems are required to operate in high-frequency such as 13.56 MHz in order to achieve a high power
density of transmission coils. Thus the AC-DC converter in the receiving side is demanded to operate
at high-frequency. In such high-frequency region, the reflected power occurs when the input
impedance is not matched to the characteristic impedance of the transmission line. In other words, theinput impedance of the AC-DC converter needs to have the same impedance to the characteristic
impedance of the transmission line. In order to overcome the problem, the AC-DC converter with an
input impedance matching is proposed in this paper.
The proposed AD-DC converter achieves the input impedance matching with a simple circuit
configuration. It means that the converter can obtain sinusoidal input current and unity input power
factor without a high-frequency switching except the diodes. In this paper, the impedance matching
characteristics and the analysis of the operational modes of the proposed circuit are presented. The
experimental results confirmed that the proposed converter enables a conversion from 13.56-MHz AC
to DC with the sinusoidal input current. In this operation condition, the input impedance is 29.6 +j0.51 . Because the design value of the input impedance is 50 + j0 , there is a non-negligible error
on the real part. This is attributed to the parasitic capacitances on the diodes. In order to solve thisproblem, an improved AC-DC converter is proposed newly. From experimental results, it achieves the
input impedance of 52.7 – j0.02  Besides, the reflection coefficient is suppressed by up to 94.5%
compared with that of the conventional capacitor input-type diode bridge rectifier (CI-DBR).

I. Introduction
In recent years, wireless power transfer methods have been attracted in the community [1-2]. In
particular, the wireless power transfer with magnetic resonance coupling (MRC) which is reported by
A. Karalis et al. in 2007 is heavily studied [3-4]. The MRC shows better features compared with theconventional wireless power transfer methods such as an electromagnetic induction and a microwave
transmission. First, the MRC allows a wireless power transfer in a middle-range transmission distancesuch as a few dozen of centimeters at high efficiency of over 90% [5]. Such high transmission
efficiency cannot be achieved with an electromagnetic induction. On the other hand, the transmission
efficiency of a microwave transmission is greater than the others. However, the conversion efficiency
between microwaves from electrical power is mere 84.4% [6].

Second, the declination of the transmission efficiency caused by a position gap of the transmitting
devices is relatively small. These advantages are featured by the characteristic constructions of thetransmitting coils. The transmission efficiency at a resonance frequency is expressed by (1). It
represents the transmission efficiency is determined by an only product of the quality factor of thetransmitting coil Q and the coupling coefficient k. The transmitting coils have a high quality factor Q
owing to the low parasitic resistance of the coils. The high quality factor allows an efficient wirelesspower transfer in a middle-range transmission distance even when the coupling coefficient is small [78].
1(1)
1  2 kQ
Considering to apply the wireless power transfer with MRC into the battery chargers for electricalvehicles (EVs), the coils for the transmitting side and receiving side are planted on the ground inparking areas, and underneath of EVs, respectively. This technology brings convenience to users
because users are not required to charge the battery with electrical cables [9].
In the wireless power transfer system with the MRC, the size of the transmitting coils depends on
the transmitting frequency [10]. Thus, the wireless power transfer systems are expected to operate inhigh-frequency in order to achieve a high power density. Moreover, MRC should be operated inindustrial scientific medical (ISM) band such as 13.56 MHz and 27.12 MHz because noise from thewireless power transfer systems is prohibited to influence the operation of neither the electronic
devices nor the radio communication equipment.
On the receiving side of wireless power transfer systems, the AC-DC converter, which converts thepower from 13.56 MHz to DC, is necessary. The previous studies have reported that the use of thediode bridge rectifier on the receiving side [11]. The diode bridge rectifier has a distorted input current
and low input power factor. Furthermore a diode bridge rectifier is connected to a DC load, the input
impedance of the diode bridge rectifier depends on a load on a DC side. In such high-frequency region,
an impedance matching is very important in order to suppress the reflected power [12]. Eventually, thereflected power which is generated by the input impedance mismatching decreases the transmission
efficiency [11]. Besides, the characteristic impedance of the transmission line such as a coaxial cable
is set to 50 + j0 . Although, the matching function is required in the AC-DC converter, a method of
an input impedance matching for the AC-DC converters has not been reported.
This paper proposes and demonstrates an input impedance of the AC-DC converter which can bematched to the characteristic impedance of 50 + j0 . First, the configuration of the proposed AC-DC
converter is described. The AC-DC converter achieves the input impedance matching without highfrequency switching devices except the diodes. Then, the operation modes of the proposed converter
are analyzed. In particular, the formulas for the input current are derived. Finally, the AC-DC
converter is tested experimentally.

II. Proposed AC-DC Converter with Input Impedance Matching
2.1. Requirements for Input Impedance Matching
Generally, high-frequency circuits are constructed with an impedance matching in order to suppress
reflected power [12]. The impedance matching can be described as; the output impedance of the power
supply and the input impedance have equal impedances to the characteristic impedance of thetransmission line. In particular, the characteristic impedance of a 50  is used widely. For this reason,
50  is used as the characteristic impedance and also the input impedance of the AC-DC converter inthis paper. Besides, the input impedance of AC-DC converters with a battery as a load is required tomatch an input impedance regardless of the load conditions.
In general, the characteristic impedance of transmission lines, which is the reference value of the
input impedance of the AC-DC converter, does not include an imaginary part. Thus, an input voltage
and current of the AC-DC converter are required to fulfill the following requirements, where Żin is the
input impedance, Vin is the fundamental input voltage of the AC-DC converter, İin is the fundamental
input current and  is the phase angle between the input voltage and the input current.
(a) Z in  Vin Iin  50 (b) Input power factor is 1 (cos  = 1)
In a low-frequency region, power factor correction (PFC) circuits with a PWM control are usedwidely [13]. The PFC circuits with a PWM can satisfy the above-mentioned conditions easily due tothe input current control with the PWM in a low-frequency region. However, the switching frequency
for the PWM requires higher frequency than an input frequency. Thus, it is difficult to operate theconventional PFC circuits when the input frequency is constrained in a high-frequency such as 13.56
MHz. In conclusion, an AC-DC converter with a simple circuit configuration which can achieve theinput impedance matching is required on the receiving side of the wireless power transfer system.

2.2. Circuit configuration
Fig. 1 presents the circuit configuration of the proposed AC-DC converter. The proposed converter
consists of the resonant-type rectifier which is reported by K. Matsui et al. [14] and the bidirectional
boost chopper. The resonant-type rectifier achieves a PFC operation using a resonance between the
inductor which is connected in series to the input terminal and the capacitors in parallel to the upperarm. The resonant-type rectifier has been demonstrated in a commercial frequency in [14]. However,this converter causes a low power density in the low-frequency operation because a bulky inductor andcapacitors as resonance components are required. Additionally, the possibility of the input impedance
matching is not discussed. In this paper, the resonant-type rectifier is operated at high-frequency.
Furthermore the function of the input impedance matching is evaluated. The high-frequency operation
improves the power density owing to the downsizing of the passive components.
In addition, Ref. [14] pointed out that the amplitude of the input current and the input power factor
hinge upon a load condition when a resistance load is connected directly to the resonant-type rectifier.
It means that the input impedance of the stand-alone resonant-type rectifier depends on the loadconditions. In order to overcome this problem, the bidirectional boost chopper is connected at theoutput side of the resonant-type rectifier in Fig. 1. The bidirectional boost chopper is operated inpurpose to fix the operating point which is decided by the rectifier output voltage vch of the resonanttype rectifier. The MOSFET S1 is used for an initial charge of the C4 instead of a diode because theinput power factor closes to zero when the rectifier outputs voltage vch is around zero.
The control for the chopper circuit does not need a high dynamic response. Thus, the chopper circuit
does not need either a high-speed or a high-frequency switching. Thus, the chopper may be operated ata low switching frequency such as 20 kHz. However, a switching frequency of 100 kHz is selected inthis paper with the objective of the downsizing of the inductor L2.
fsw = 100 kHz

Resonant-type rectifier
irec
Żin

D1
iin

D2
C1

C3

D3

C4

C5 +

L2

VB

S2

+

vin L1
13.56 MHz

ich
ic

C2

S1

vch

vconv

D4

Fig. 1. The proposed AC-DC converter.

2.3. Control method of bidirectional boost chopper
Fig. 2 shows the control block diagram of the bidirectional boost chopper. The voltage control iscomposed by an automatic voltage regulator (AVR) with a PI control and an automatic current
regulator (ACR) as an inner loop of the AVR, where the natural angular frequencies of the AVR andACR are 400 rad/s and 4000 rad/s respectively. Note that Tic and Tiv are the integral time of the ACR
and AVR, respectively. The input impedance is determined by the relation among the voltage ratio V,
the inductance L1 and the capacitances C1 and C2 where the voltage ratio is the ratio of the rectifier
output voltage vch to the input maximum voltage Vm. The voltage ratio should be stabilized in constantin order to obtain the intended input impedance when the input power is constant. Thus, the voltage
ratio is controlled through the rectifier output voltage control, where the reference value of the rectifier
output voltage is obtained by (2).
*vch
 Vm  V*(2)
In the proposed circuit, a fast dynamic response of the bidirectional boost chopper is not necessary
because the input maximum voltage is not changed fast. For this reason, an inexpensive generalpurpose controller can be used. Note that, the capacitance C3 is negligible in the AVR because thecapacitance C3 is enough smaller than C4 in the control block diagram.
vch

vch*

V*

1
+
1  sTiv

ich*

PI

1
+
1  sTic

+

PI

vconv

irec

1
sL2

+
ich

+

1
sC4

vch

Vm

Fig. 2. The control block diagram for the proposed AC-DC converter.

III. Operation modes of the proposed AC-DC converter
Fig. 3 shows the operation modes of the proposed converter. Note that the bidirectional boost
chopper with AVR is illustrated as an ideal DC voltage source.
Fig. 4 illustrates the simplified operation waveforms when the proposed converter is operated at theunity input power factor. In this chapter, the forward voltage drop of the diodes is ignored forsimplicity. The circuit operations in each four operation are described in the following statements.

3.1. Operation mode I
Input current iin flows through the inductor L1, capacitors C1 and C2. The capacitors C1 and C2 aredischarged and charged respectively by the input current. The input current in a mode I iin_I is derived
from a circuit equation as  2 CV m
 Cvcos1t  cosin t   1 ch sin 1t
iin _ I t   in 2 1(3),2
22 in  1 
where C=C1=C2, L is the inductance of the inductor L1, Vm is the maximum input voltage, in is the
input angular frequency and 1 is the resonance angular frequency which is expressed as (4). Note thatthe resonance angular frequency is obtained as a series resonance of the inductance L and the twocapacitances C.
1
1 
(4)
LC 2
I

II

III

IV

vin
irec
D 1 C1vcu

iin L1 idu icu

irec
D1 C1

D 2 C2vcv

idv icv

vch

vin

D

C

vcu 2 2
icu idv icv

vin

C3
D3

iin L1 idu

D4

iin L1 idu icu

D 2 C2idv icv

vch
C3

D4

Mode III

D1 C1

vcv

iin L1 idu

vcu

irec

0

vcv
icv v
ch

0

icu idv

C3

0

vcu
vcv
idv

idu

D2 C2

vin
D3

0
vch

Mode II

vin
D3

vch

D4

irec
vcu

iin

vcv
C3

D3

Mode I
D 1 C10

D4Mode IV

Fig. 3. Operation modes of the proposed circuit.

icu
icv
TI
T

Fig. 4. Simplified waveforms of the proposed
AC-DC converter.

From (3), the input current includes both the frequency components; resonance angular frequency

1 and input angular frequency in. Thereby, the input current is not a complete sinusoidal waveform.
The capacitor C1, which had charged in operation mode III, is discharged by the input current. In
contrast, the capacitor C2 is charged gradually. After the capacitor voltage vcv reaches to the DC
voltage vch, the next operation mode will start.
Incidentally, the period of the operation mode I TI is equal to the discharging time of the capacitor
C1. Thus, this period TI can be derived from (5) with the numerical analytical approach.
vch1  cos1TI   21Vm 2 1 sin inTI  in sin 1TI   0(5)
2
2in  1 

3.2. Operation mode II
The input current commutates to the diodes D1 from the capacitor C1 because the diodes D1 and D4
become a turn-on. The input current flows into the path that is formed by inductor L1, diodes D1 andD4. In this mode, the smoothing capacitor is charged by the input current, which is expressed byv
iin _ II t   iin _ II cos2 t  TI   ch sin 2 t  TI 
t TI
2 L
V cosinTI
 m2
(6),
in  22 L in cos2 t  TI   in cosin t  TI 
V sin inTI
 m2
in  22 L in sin in t  TI   2 sin 2 t  TI 
where iin_II(TI) is the initial value of the input current which is also expressed as iin_I(TI), 2 is theresonance angular frequency which is expressed by (7). Also, the input current in the mode II is not acomplete sinusoidal.
1
2 
(7).
LC 3
Besides, (6) can be simplified as (8), when the resonance angular frequency 2 is enough small tobe ignored. Generally, the capacitance C4 has a large capacitance because it should compensate thevoltage fluctuation which is caused by the bidirectional boost chopper.
v
V
iin _ II t   iin _ II t TI  ch t  TI   m cosinTI  cosint 
(8)
L
in L
Besides, the capacitor voltages vcu and vcv are not changed in this mode. The mode II continues until
the input voltage polarity is changed from positive to negative.

3.3. Operation mode III
In this mode, the input current flows in the opposite direction to the mode I via C2, C1 and L1. In thismode, the input current is expressed by iin_III(t) = -iin_I(t-T/2) where T is the input period. The capacitors
C2 and C1 are discharged and charged respectively by the input current. The mode III continues until
the capacitor voltage vcu reaches to the DC voltage vch. The period of the mode III is same to the one ofmode II.

3.4. Operation mode IV
The input current commutates to the diodes D2 and D3 from the capacitors C1 and C2. The input
current is expressed by iin_IV(t) = -iin_II(t-T/2).
The amplitude of the input current and the input power factor are determined by the parameters thatare used for the resonance; capacitors C1, C2 and inductor L1 and voltage ratio V. Hence the intended
input impedance of the AC-DC converter can be obtained by a designing these parameters properly.
However, the design method is omitted in this paper because of space limitation.

IV. Simulation Results of the Proposed AC-DC Converter
In this chapter, the simulation result of the proposed AC-DC converter is shown. Table I presents
the simulation conditions. In the simulation, the rated power is assumed as 1 kVA because the wirelesspower transmission systems are expected to apply to the EV charger.
Fig. 5 presents the simulation waveforms of the proposed AC-DC converter. It is confirmed that therectifier output voltage vch and chopper current ich track to the each reference value. Thus, thebidirectional boost chopper is operated normally according to the PI control. Moreover, a sinusoidal
input current and unity input power factor are achieved. In this simulation, the input impedance iscalculated from the fundamental component of the input voltage and input current as 50 + j0.0  withusing (9), where Vin_1st is the fundamental component of the input voltage, Iin_1st is the fundamental
component of the input current and  is the phase angle between the input voltage and input current of
the fundamental component.
Vin _1st
Vin _1st
Zin 
cos  jsin (9)I in _1st
I in _1st
It means that the proposed converter is capable of the input impedance matching to the 50 + j0  at
a reflection coefficient of 0.0% which is defined byZ  Zin
PR
 0



(10),
Z 0  Z in
PF
where Ż0 is the characteristic impedance, PF is supplied travelling power to the AC-DC converter, PR
is the reflected power which occurs at the input terminal of the AC-DC converter. Note that thereflection coefficient is used widely in a research field of high-frequency circuits to evaluate thecircuit performance. The squared reflection coefficient means the ratio of reflected power to thetraveling power. In this paper, the input impedance is matched to a 50 + j0  because it is mostcommon impedance, although the input impedance of the proposed circuit can be matched to otherintended impedances.
400

Table I: Simulacion conditions.
Items
Inductors

Capacitors

Value
L1

551 nH

Input voltage vin

0-400
Input current iin
10
0
-10

L2

2.3 mH

C1, C2138 pFC3

940 nFC4

0 mF

250.0
349.8
Chopper current ich
-2.4

C5

220 mF

-2.8

350.2

Chopper voltage vch-3.2 0

vch
vch*

ich
ich*

0.1

0.2
Time t [msec]

0.3

Fig. 5. Simulation result of the proposed AC-DC
converter.

V. Experimental Results of the Proposed AC-DC Converter
Experimental verifications are shown in this chapter. Table II provides the circuit parameters for theexperimental setup. Note that, the silicon carbide schottky barrier diodes (SiC-SBDs) are used in therectifier because rectifying diodes are required to have a performance to rectify the 13.56–MHz AC.
Besides, the design method of the resonance parameters, which affects the input impedance, is omitted
in this paper due to the page limitation.

Table II: Parameters of the circuit components.
Items
MOSFET

S1 , S2

Diode

D1-D4
L1

Manufactures

Model number

Value

VishayIRFB11N50APBF

500 V, 11 A

Cree

C3D08060A

600 V, 8 A

TDK

VLF10040T-1R5N8R9 (Remodeled)

950 nH

Inductor
L2--

2.3 mH

C1, C2

TDK

C3216C0G2J151JT

150 pFC3TDKCKG57NX7R2J474M

470 nF(in parallel)

C4nichicon

UPW2V221MRD

220 mF(in parallel)

C5

BHC Components

ALS30A221DB450

220 mF

Capacitor

5.1. Fundamental characteristics
Fig. 6 shows the experimental waveforms of the proposed AC-DC converter. It is clear that power
conversion from 13.56 MHz to DC is achieved by the proposed converter with the sinusoidal input
current and unity input power factor.
Fig. 7 presents the harmonics analysis results of the input voltage and current which is conducted inorder to derive the input impedance from the experimental waveforms shown in Fig. 6. Note that, theprobes; a differential probe (Tektronix, P5205) and a current probe (Tektronix, TCP312), which areused in these experiments provide a limitation to the frequency bandwidth at 100 MHz. For this reason,
the harmonics components over 7th are considered as reference values.
From the calculation with (9), the input impedance is calculated as 29.6 + j0.51 . The input
impedance includes the non-negligible error on the real part. This is attributed to the parasitic
capacitances of the SiC-SBDs. In the next subsection, the proposed circuit is improved in order toachieve intended input impedance.
Fig. 8 shows the reason of the error on the input impedance. In a high frequency region, theparasitic capacitances Cp1-4 of the diodes cannot be ignored. Especially, an effect of the parasitic
capacitance increases when a reverse voltage of the diodes is high. The parasitic capacitances make aleakage current path during the operation modes I and III. Owing to the leakage current path, the input
current is separated into the primary current path and the leakage current path. Thus the relation
between the resonance capacitance C and the input impedance is different from the intended one. The
combined capacitance on the upper arm is composed by the parasitic capacitances and additional
capacitors. By contrast, the lower arm has only parasitic capacitances. Thus the current flowing
through the upper and lower arms becomes unbalanced. It means that the design procedure of the
Input voltage [p.u.]

1

Input voltage vin [50V/div]0
Input current iin [2A/div]

10-2
10-3
1

Input current [p.u.]

0
Rectifier output voltage vch [50V/div]0
Output voltage VB [100V/div]

0

10-1

10-1
10-2
10-3
1

40[nsec/div]

Fig. 6. Experimental waveforms of the proposed
AC-DC converter.

3

5

7

9 11 13 15 17 19

Number of harmonics

Fig. 7. Harmonics analysis of the input voltage
and input current of the proposed AC-DC
converter.

resonance capacitance C is complicated. In order to solve the above-mentioned problem, the improved
AC-DC converter is newly proposed in the next subsection.

5.2. Improvement of impedance matching characteristics
Fig. 9 shows the improved AC-DC converter with an input impedance matching. Additionally,
Table III provides the circuit parameters for the experimental setup with the improved AC-DC
converter. The two resonance capacitors; C6 and C7 are added into the lower arm in the purpose ofbalancing the impedances between the upper arm and lower arm. Owing to these additional capacitors,
the input impedance can be designed by considering the parasitic capacitances easily because the input
current is shunted equally to the each arm in the mode I.
Fig. 10 presents the operation modes of the improved AC-DC converter. The operation modes of
the improved AC-DC converter are similar to the one of the AC-DC converter without an
improvement. The difference between the Fig. 3 and Fig. 10 is the resonance current path in the modes
I and III. The input current is divided to the path through an upper arm (L1, C1, C2) and the paththrough a lower arm (L1, C6, C7). For this reason, the relationship between the circuit parameters andinput impedance has changed. In this paper, the parasitic capacitances of the diodes are used as thecapacitors C1, C2, C6 and C7. Thus, the resonance inductance is changed from 950 nH to 1.5 mH.
Fig. 11 presents the operation waveforms when the parasitic capacitance is used as a resonancecapacitor. The DC output is obtained from an 13.56-MHz AC. Additionally, it is clarified that thesinusoidal input current and unity input power factor are obtained.
Fig. 12 shows the harmonic analysis results with the improved AC-DC converter. The input current
total harmonic distortion (THD) of 8.1% is achieved with the bandwidth up to 20th. A THD with the
improvement is suppressed by 27.7% compared with the one without improvement.
In addition, the input impedance is calculated
irec
D1 C1 Cp1
icu

iin L
1

D2 C2 Cp2
vcu
icv

S1

vcv

irec
vch

vin
Cp3

D1

Fig. 8. Effect of the parasitic capacitances in themode I.

C1

Diode

S1 ,
S2
D1D4

C6

D3

D4

Model number

Value

VishayIRFB11N50APBF

500 V, 11 A

C3D08060A

L1

TDK

VLF10040T1R5N8R9

1.5 mHL2--

2.3 mH(34.3 pF)

vcu

vcv

idv icv

vch

C6

D4

C3

TDK

CKG57NX7R2J474M

C4nichicon

UPW2V221MRD

220 mF(in parallel)

C5

BHC
Components

ALS30A221DB450

220 mF

iin L1 idu

D2 C2

vcu
icu idv

vin

C3
D3

C7

C6

D4

vcu

D 2 C2

irec

vch
C3

D3

D4

Mode III

D1 C1

vcv

idv icv

C6

C7

Mode II
irec

iin L1 idu icu

vcv
icv v
ch
C3

D3

Mode Ivin
470 nF(in parallel)

irec
D1 C1

D 2 C2

vin

600 V, 8 A(Parasitic capacitancesof the diodes)

Capacitor

vconv

C7
D4

iin L1 idu icu

D 1 C1

-

C4 +

irec

Cree

C1-2,
C6-7

VB

S2

Fig. 9. Improved AC-DC converter with input
impedance matching.

D 1 C1

Manufactures

Inductor

C2
C3

C5 +

L2

vch

13.56 MHz

Table III: Parameters of the circuitcomponents in the improved AC-DC
converter.
MOSFETic

vin L1

D4 Cp4

Leakage current path

Items

D2

iin

C3
D3

Żin

ich

C7

iin L1 idu

D2 C2

vcu
icu idv

vin
D3

C6

D4

vcv
icv v
ch
C3
C7

Mode IV

Fig. 10. Operation modes of the improved ACDC converter.

Input voltage [p.u.]

1

Input voltage vin [50V/div]0
Input current iin [1A/div]

10-1
10-2
10-3
1

Input current [p.u.]

0
Rectifier output voltage vch [50V/div]00Chopper current ich [0.5A/div]

10-1
10-2
10-3

40[nsec/div]

1

3

5

7

9 11 13 15 17 19

Number of harmonics

Fig. 11. Operation waveforms of the improved
AC-DC converter.

Fig. 12. Harmonics components of the input
voltage and input current in the improved ACDC converter.

as 52.7 - j0.02 . The reflection coefficient, which is calculated by (9), is  = 2.6%. The reflection
coefficient is used to evaluate a matching characteristic of the high-frequency circuit in the research
field related to high-frequency circuit. If the impedance matching is completely held, the reflection
coefficient is zero.
It means that the ratio of the reflected power which occurs at the input of the circuit to the travelling
power is lesser than 0.1%. It is clarified that the proposed AC-DC converter suppresses the reflected
power from the above experiments. Thus, the proposed circuit can be applied to the receiving side of
the wireless power transfer system.

5.3. Comparison of the reflection coefficient

VI. Conclusion

Reflection coefficient  [%]In this subsection, the reflection coefficients are compared among the conventional CI-DBR,
proposed converter without the improvement and the proposed converter with the improvement.
Generally, the input impedance matching of the CI-DBR is not achieved in the reason of the
distorted input current and low input power factor. Moreover, the reflection coefficient of theconventional CI-DBR depends on the load conditions and a smoothing capacitor widely. In contrast,
the reflection coefficient of the proposed circuit does not depend on the load condition and a value ofthe smoothing capacitor.
Fig. 13 shows the comparison result of the reflection coefficient which is calculated from (10)among the conventional CI-DBR and the proposed AC-DC converters. Note that the conventional CIDBR has a smoothing capacitor of 0.47 mF. The proposed converter with the improvement and one
without the improvement achieve the input impedance matching with the reflection coefficients of
2.6% and 29.3% respectively. It is clear that the reflection coefficient can be suppressed compared tothe conventional CI-DBR. In particular, thereflection coefficient with the improved AC-DC
50
Without improvement(Fig. 1)converter suppresses the reflection coefficient by
40
With improvement
94.5 % compared with the conventional CI-DBR(Fig. 9)30
with a load of 25 .
20
10
0

25  33.3  50  100  Proposed circuit
This paper discussed the AC-DC converter
Conventional CI-DBR
which converts power from 13.56-MHz AC to
DC for a receiving side of a wireless power Fig. 13. Comparison of the reflection coefficienttransfer system with a sinusoidal input current. between the conventional CI-DBR and proposed
The wireless power transfer systems are required circuit.

to operate in high-frequency such as 13.56 MHz in order to achieve a high power density oftransmitting coils. Thus, the AC-DC converter in the receiving side is required to operate at highfrequency. In such high-frequency region, the reflected power occurs when the input impedance is notmatched to the characteristic impedance of the transmission line. In other words, the input impedanceof the AC-DC converter needs to have the same impedance to the characteristic impedance oftransmission lines.
In order to overcome the above-mentioned problem, two AC-DC converters are proposed andexperimentally tested. The both AC-DC converters achieve the input impedance matching withouthigh-frequency switching devices except the diodes with a simple configuration. The experimental
results confirmed that the proposed converter with the improvement enable a conversion from 13.56MHz AC to DC with the sinusoidal input current. From the experimental results, one of the proposed
converters has an error on the real part of the input impedance. In order to improve this problem, thesecond converter is proposed. The proposed converter has two additional capacitors compared to the
AC-DC converter without the improvement. The experimental results confirmed that the input
impedance of 52.7 - j0.02  is achieved. It means that the reflection coefficient is suppressed by up to
94.5 % in comparison with the conventional diode bridge rectifier with a load of 25 .

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