# Inductive Current Sensing for Closely Spaced Antiparallel Conductor Pairs

Dirk Rudolph, Lars Lindenmüller, Steffen Bernet

► To cite this version:

Dirk Rudolph, Lars Lindenmüller, Steffen Bernet. Inductive Current Sensing for Closely Spaced Antiparallel Conductor Pairs. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0297. hal-05080469

HAL Id: hal-05080469

https://utc.hal.science/hal-05080469v1

Submitted on 22 May 2025

HAL is a multi-disciplinary open access
archive for the deposit and dissemination of sci-
entific research documents, whether they are pub-
lished or not. The documents may come from
teaching and research institutions in France or
abroad, or from public or private research centers.

L'archive ouverte pluridisciplinaire HAL, est
destinée au dépôt et à la diffusion de documents
scientifiques de niveau recherche, publiés ou non,
émanant des établissements d'enseignement et de
recherche français ou étrangers, des laboratoires
publics ou privés.

<figure><img src="image_1.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
# Inductive Current Sensing for Closely Spaced Antiparallel Conductor Pairs

Dirk Rudolph

TUD Dresden University of
Technology

Chair of Power Electronics
dirk.rudolph@tu-dresden.de

Lars Lindenmüller
F&S

PROZESSAUTOMATION
GmbH

lars.lindenmueller@fs-aut.de

Steffen Bernet

TUD Dresden University of
Technology

Chair of Power Electronics
steffen.bernet@tu-dresden.de

## ACKNOWLEDGMENT

We would like to thank ABB for funding this research. In particular, we acknowledge the support of Matti Laitinen (ABB Finland) and Sami Pettersson (ABB Corporate Research, Switzerland) for their valuable input and collaboration.

**Index Terms**—Sensor, Current sensor, Integrated Rogowski coils, Fast fault detection, Component for measurements

## Abstract—

Based on two individual current measurements of two antiparallel conductors (e.g., the DC terminals of an IGBT module), a method is described that relies on a common current measurement. This approach avoids placing a sensor between the two conductors, thus not limiting the isolation distances between them and enables current measurement even where there is no space for an inductive sensor between the two conductors.

## I. INTRODUCTION

In many applications using power electronic modules (e.g., IGBT modules) like inverters, drives, and DC/DC converters, it is essential to measure the current through the switching devices or capacitors. This is crucial for circuit control and protection of the semiconductor devices. To achieve optimal switching behavior, connections with low leakage inductance are generally placed close together (e.g., power electronic modules, power capacitors, intermediate circuits, etc.).

In these applications, the stray inductance in the commutation loop ranges from several tens of nanohenry for low-voltage applications [1] and several hundreds of nanohenry for medium-voltage applications [2]. This is significantly smaller than the load inductance, leading to

rapid current changes during commutation compared to the more stable current during the on-time of the power semiconductor. A typical current waveform is shown in Fig. 1.

<figure><img src="image_2.png" /><figcaption>Fig. 1. Schematic of a Commutation Cell and Simplified Current Waveforms</figcaption></figure>

The currents in the commutation loop can be divided into two main categories with the following characteristics:

### Quasi Static Current

* One device conducts the load current, the other one not ($i_{T1} = -i_L$, $i_{T2} = 0$ A or $i_{T2} = i_L$, $i_{T1} = 0$ A)

* Small rates of current rise ($\frac{di_L}{dt} = \frac{V_{DC}}{L}$)

### Pulse Current (gray highlighted)

* Current commutates from the IGBT to the Freewheeling diode (FWD) and vice versa

* Same current waveform but shifted by the load current ($i_{T2} = i_{T1} + i_L$)

* Same rise in current ($\frac{di_{T1}}{dt} = \frac{di_{T2}}{dt}$)

* High rates of current rise (depends on the switching characteristics of the semiconductors, the stray inductance in the commutation loop and the DC-link voltage)

For current control, a high upper bandwidth is not required, as measuring the quasi-static current is sufficient. Given the known switching signals and zero current
periods, the quasi-static current can be reconstructed from the pulse currents, allowing measurements of the second category to be used for control.

For safe operation, fast short-circuit detection is essential. This can be achieved by the widely used desaturation detection or by measuring and evaluating the pulse current. Additionally, dynamic current balancing necessitates a high-bandwidth current measurement.

## II. STATE OF THE ART

State of the art current sensing typically uses Hall effect sensors, available in a wide range of products. However, these devices have limited bandwidth, require significant installation space, and can be expensive. Newer approaches involve higher integration, such as placing a small Hall sensor inside an IC near a bus bar or PCB conductor track [3]. This solution is more cost-effective and compact but still lacks high bandwidth.

Another method involves shunt-based current sensors [4], which offer high bandwidth and can measure DC currents. However, these sensors must be inserted into the load current path, preventing galvanic isolation and increasing both inductance and resistance, leading to higher losses.

A third approach is the use of inductive sensors, particularly Rogowski coils. These sensors provide a wide bandwidth and are easy to manufacture. Their main drawback, compared to Hall effect sensors, is the limited lower bandwidth. However, unlike shunt-based sensors, Rogowski coils can achieve galvanic isolation. An overview of proposed Rogowski coil designs and their characteristics is provided in [5].

Typically, two Rogowski coils are designed to measure the pulse current of an IGBT module. Exemplary recent designs can be found in [6], [7], [8] and [9].

Considering $\frac{di_{T1}}{dt} = \frac{di_{T2}}{dt}$ during commutation, it is possible to connect both coils, $M_{R1}$ and $M_{R2}$, in series [10]. This configuration doubles the signal amplitude and requires only one sensor evaluation circuit, such as an integrator (see Fig. 2). The two adjacent antiparallel conductors, carrying pulse currents of the same amplitude and in forward and reverse directions, are referred to as a double conductor in the following sections. This can be either the connectors of the capacitor or the DC-link connectors of the power modules.

Another approach is the use of coils which are not necessarily shaped as a closed loop for example described in [11] and [12]. Here, one coil is placed between the DC-link bus bars respectively the capacitor terminals.

<figure><img src="image_3.png" /><figcaption>Fig. 2. Schematic and Implementation of Two Series-Connected Rogowski Coils for Current Measurement in a Double Conductor</figcaption></figure>

All the described methods limit the isolation distance because the whole coil or parts are located between both DC-link potentials. In the following a solution is described, which allows an accurate current measurement without restriction of the isolation distance.

## III. CURRENT MEASUREMENT BASED ON OPEN LOOPS

The new concept is based on the series connection of two closed coils surrounding a double conductor — as seen in the terminals of an IGBT module — but without the inner section.

### A. Concept

By omitting the inner section between the double conductor, two series-connected open coils are formed, which do not restrict the insulation distance between the conductors or even can be used if there is less or no space available between the connection (see Fig. 3).

<figure><img src="image_4.png" /><figcaption>Fig. 3. Resulting Current Measurement with Two Half-Open Inductive Sensors</figcaption></figure>
The resulting sensing loop will be less sensitive to enclosed double conductor (internal fields) and will also sense unintended external fields. Following chapters will investigate the resulting behavior.

## B. Internal Magnetic Field

As shown in Fig. 3, no windings are placed between the two conductors, but the magnetic field is the strongest in this region, which would result in a significant measurement signal. Consequently, the proposed measurement coil is expected to exhibit lower sensitivity compared to the classical solution with a sensor in the inner region between the conductors, as depicted in Fig. 2.

Due to the change of the winding direction, inner magnetic field caused by a changing current will induce a voltage which will have the same polarity in every part of the coil (see Fig. 4).

<figure><img src="image_5.png" /><figcaption>Fig. 4. Magnetic Field Generated by an Internal Double Conductor and the Induced Voltage in the Rogowski Coil</figcaption></figure>

## C. External Magnetic Field

Due to the design of the coil, with two "open" Rogowski coils connected in series, external magnetic fields are also sensed. Fig. 5 illustrates the induced voltage in the windings when the magnetic field is generated by an adjacent IGBT, such as Infineons EconoPACK+™ or if another module like Infineons EconoPACK™ module is positioned in y-direction. The two vertical coil parts on each side counteract partly and the horizontal elements induce a voltage with the same polarity in a series connection.

Fig. 6 shows the induced voltage, if there is an external magnetic field in x-direction, e.g. caused by an power module, which is arranged face to face to the IGBT.

Both bottom and top elements on the sides conteract partly. The bottom and top winding induce a voltage with the same polarity in this series conection.

<figure><img src="image_6.png" /><figcaption>Fig. 5. Magnetic Field of an External Double Conductor in the y-Direction and the Induced Voltage in the Rogowski Coil (e.g. coupling of another Phase)</figcaption></figure>

<figure><img src="image_7.png" /><figcaption>Fig. 6. Magnetic Field of an External Double Conductor in the x-Direction and the Induced Voltage in the Rogowski Coil</figcaption></figure>

Complete suppression of external field interference is not possible. However, due to the greater distance between the measurement coil and the external field, compared to its proximity to the internal field, the signal amplitude induced by the internal field is significantly larger than that from the external field. This results in a measurement that is more sensitive to the desired internal currents, while minimizing interference from external magnetic sources. Since the measuring method presented here detects rapid changes in the magnetic field, it is insensitive to the slowly changing magnetic fields of the phase output if the load has a high output inductance of several microhenries.

# IV. DESIGN AND TEST OF THE CURRENT SENSING CIRCUITRY

To validate the proposed measurement approach, a prototype was implemented and tested under double-pulse operation to get first measurement data which is than be used for designing the necessary integrator.
## A. Test of the Inductive Pickup Coil

Fig. 7 shows the realized prototype, the winding scheme as an overlay in the top left corner, the evaluation circuitry with an active integrator and the corresponding placement on top of the DC-link connectors of a EconoDUAL™ like module. The prototype was tested in double-pulse operation and Table I lists the utilized measurement equipment.

<figure><img src="image_8.png" /><figcaption>Fig. 7. Sensor Prototype with Highlighted Windings Structure Overlay and Evaluation Circuitry</figcaption></figure>

<table><caption>TABLE I
MEASUREMENT EQUIPMENT</caption><thead><tr><th>Measurement</th><th>Device</th><th>Description</th></tr></thead><tbody><tr><td>Oscilloscope</td><td>LeCroy<br>WaveSurfer 44Xi</td><td>400 MHz, 8<br>Bit, 5 GS/s</td></tr><tr><td>LCR Meter<br>Bottom Device Current<br>(i<sub>T2</sub>)</td><td>Hameg HM8118<br>PEM CWT15B</td><td>—<br>16 MHz</td></tr><tr><td>Induced Voltage (v<sub>m</sub>)</td><td>Passive LeCroy<br>Probe PP008</td><td>1/10,<br>500 MHz</td></tr></tbody></table>

First the coil was characterized without connecting the integrator. Fig. 8 presents the resulting measured voltage $v_m$ at the coil's connections, the bottom device current $i_{T2}$ measured by the commercial Rogowski coil PEM CWT15, and the numerical integrated voltage multiplied by the constant factor $k_{scal} = -168 \cdot 10^6$ during the turn-on of an EconoDUAL™ module. The measurements are conducted in a double-pulse operation with a switching voltage $V_{sw}$ of 600 V and a load current of 600 A. The estimated current by integration of the voltage $v_m$ was set to zero at 20 µs to avoid a DC current offset.

The voltage shows a relatively strong oscillation, which is also visible in the derived current. However, the derived current follows the measurement from the pickup coil quite well. Depending on how the factor $k_{scal}$ is chosen, this can result in overshoot or a DC offset.

<figure><img src="image_9.png" /><figcaption>Fig. 8. Measured Voltage, Estimated Current, and Comparison With a Commercial Measurement System (Double-Pulse Operation, $V_{sw}$ = 600 V, $I_L$ = 600 A).</figcaption></figure>

For a closer optimization the parameters of the coil are investigated further.

## B. Design of the measurement system

In [13] a model of a Rogowski coil is given with the damping resistor $R_m$ (see Fig. 9).

<figure><img src="image_10.png" /><figcaption>Fig. 9. Equivalent Circuit Model of a Rogowski Coil With the Damping Resistor $R_m$.</figcaption></figure>

The proposed pickup coil encloses the introduced double conductor and measures both currents, $i_{T1}$ and $i_{T2}$.

Taking into account the pulse current characteristics outlined in Chapter I, the resulting voltage $v_{ind}$ is given by:

$$v_{ind} = M_C \frac{di}{dt} \qquad (1)$$

$$= M_C \left( \frac{di_{T1}}{dt} + \frac{di_{T2}}{dt} \right) \qquad (2)$$

$$= 2 \cdot M_C \frac{di_{T2}}{dt} \qquad (3)$$

Because there was no additional damping resistor $R_m$ in the previous measurement, the oscillations between the inductance $L_C$ and parasitic capacitance $C_C$ of
the coil is only damped by the small resistance of the PCB tracks $R_C$. Therefore an additional damping resistor is necessary.

The equivalent circuit in Fig. 9 can be described by following transfer function:

$$ \frac{V_m(s)}{I_{T2}(s)} = \frac{2 \cdot M_C s}{L_C C_C s^2 + \left( \frac{L_C}{R_m} + R_C C_C \right) s + \left( 1 + \frac{R_C}{R_m} \right)} \quad (4) $$

$L_C$ and $R_C$ were measured with an LCR-meter. If no damping Resistor $R_m = \infty$ is used, $C_C$ can be estimated with the resonance frequency of the oscillation $f_R$ by

$$ C_C = \frac{1}{(2\pi f_R)^2 L_C} \quad (5) $$

$M_C$ can be estimated by transferring (4) into the time domain and $R_m \to \infty$:

$$ M_C = \frac{C_C L_C \frac{d^2 v_m}{dt^2} + R_C C_C \frac{dv_m}{dt} + v_m}{2 \cdot \frac{di_{T2}}{dt}} \quad (6) $$

To calculate the $M_C$ a rising and falling current slope of a double-pulse test is used and evaluated in the range of 20% to 80% of the load current. An exemplary turn-on waveform for estimation of $M_C$ is shown in Fig. 10.

<figure><img src="image_11.png" /><figcaption>Fig. 10. Measured Turn-On Waveform for Calculating the Mutual Inductance $M_C$ at a Load Current of 600 A.</figcaption></figure>

The values of the elements were estimated to:

$$ L_C = 2.4 \mu H \quad (7) $$

$$ R_C = 5.3 \Omega \quad (8) $$

$$ C_C = 84.3 pF \quad (9) $$

$$ M_C = -3.15 nH \quad (10) $$

$R_m = 180 \Omega$ has been chosen to damp the oscillations after step responses.

<figure><img src="image_12.png" /><figcaption>Fig. 11. Bode Plot of the Transfer Function Assuming an Ideal Integrator.</figcaption></figure>

With the transfer function (4) and an ideal integrator

$$ \frac{V_I(s)}{V_m(s)} = \frac{1}{s} \quad (11) $$

a bode plot was created which is shown in Fig. 11.

The plot shows a cutoff frequency of approximately 12.5 MHz. In comparison, the PEM Rogowski coil of type CWT15 employed here has a maximum bandwidth of 16.0 MHz. This relatively low cutoff frequency of the pickup coil is attributed to the high values of $C_C$ and $L_C$ resulting from a large number of turns.

### C. Possible Application

The proposed sensor concept is suitable for power electronics applications that require maintaining isolation distances or where space between the conductors is insufficient for conventional sensors. A key example is current measurement in power semiconductor modules with internal busbars extending to the substrate, such as XHP2™, or in modules like XHP3™, where an internal insolation structure already exists between the DC-link connections.

This approach enables precise current measurement in power semiconductor modules and capacitors, allowing for accurate monitoring of switching events and conduction states. For instance, the current direction can be detected, enabling a MOSFET with a freewheeling diode conducting in reverse direction to be switched on while a reverse-conducting IGBT (RC-IGBT) is switched off [14], thereby reducing conduction losses.

Additionally, the sensor can be used for fast fault detection, such as identifying short circuits and overcurrent conditions, ensuring system protection.

A dedicated sensing circuit in each module enables current imbalance detection and compensation by adjusting control signals such as gate current or voltage. This
ensures even current distribution and prevents thermal
overstress of individual components.

Furthermore, the proposed concept is suitable for the
development of low-cost converters, eliminating the need
for additional current sensors while maintaining reliable
performance.

V. EXPERIMENTAL INVESTIGATION OF CURRENT
SENSING PROTOTYPE

The proposed sensor is tested in double-pulse opera-
tion and the hardware implemented non-inverting active
integrator is used for current estimation. An additional
resistor discharges the capacitor of the integrator with a
high time constant to prevent the accumulation of voltage
offsets. Additionally the current offset is removed at 2 µs
and 20 µs via post processing. This can later be done in
hardware, as the switching times will be defined using
the control signals and the sensor need to be integrated
on the gate drive unit.

Fig. 12 shows the prototype in the double-pulse
setup with an EconoDUAL™ module. The original Gate
Drive Unit is lifted by some millimeters so that the
prototype can be placed under the Driver. In the practical
application, the sensor can be integrated into the gate
driver.

<figure><img src="image_13.png" /><figcaption>Fig. 12. Sensor Prototype with Integrator inside the Double Pulse Testbench.</figcaption></figure>

Following the sensor is investigated by comparing the
commercial Rogowski coil PEM CWT15 with the output
of the integrator multiplied by the constant scaling factor
of 373 Α/V.

A. Internal Fields — Measurement of the Enclosed Double Connector

Internal fields are sensed as illustrated in Fig. 12
by enclose the DC-link terminals to the power semi-
conductor module. The resulting waveforms are schown

<figure><img src="image_14.png" /><figcaption>Fig. 13. Current Measurements of the DC-Link Connectors of an IGBT Module during Double-Pulse Operation at a Switching Voltage of 600 V and various Load Currents.</figcaption></figure>

in Fig. 13 for a turn-off and turn-on of the IGBT with
different currents and a DC-link voltage fo 600 A

The measurement waveforms match quite well the
measurement of the commercial measurement system
except a small overshoot during the reverse recovery
current peak. This can be explained by the frequency
response in Fig. 11 and the overshoot of the magnitude
at 10 MHz.

B. External fields — Measurement of the Unintentional
External Double Connectors

A switching event of an external double connector
is reenacted by shifting the sensing circuit by 50 mm
in the y-direction, corresponding to the distance of a
neighboring phase inside of an EconPACK+™ module.
For the test setup, this means that no conductor is
enclosed in the coil. The IGBT is operated in double-
pulse mode and the degree of unwanted coupling into
the sensor is measured (Fig. 14).

External fields are also sensed, but their magnitude
depends on location and remains significantly smaller
than the fields of the internal fields. In this setup,
the measured current amplitude of the external field is
approximately 8% of the internal field amplitude.

C. Benchmark Criteria

An important design criterion for such sensors is
maximizing sensitivity to internal fields while minimiz-
ing external field interference.

For that an arrangement of a 6-Pack (e.g.
EconoPACK™) is assumed. The relationship between
<figure><img src="image_15.png" /><figcaption>Fig. 14. Measurement of Unwanted Coupling by Shifting the Sensor 50 mm in the y-Direction. The commercial coil remains connected to the IGBT module.</figcaption></figure>

the mutual inductance of external to internal fields is defined as a criterion:

$$B_{6=\text{Pack}} = \left| \frac{M_{\text{int}}}{M_{\text{ext,6-Pack}}} \right| \quad (12)$$

The higher the value the more robust is the measurement against a field which is caused for this criteria by an adjacent IGBT of another phase in a 6-Pack in form of a EconoPACK+™ module

The mutual inductance of an adjacent EconoPACK+™ to the sensing coil was simulated which results in a $B_{6-Pack}$ value 6.9.

Since the resulting magnetic field of the double conductor is known, the coil can be optimized according to the resulting magnetic field lines. For this purpose, the windings can be adjusted accordingly and the $B_{6-Pack}$ value increased. Fig. 15 shows a scheme were the turns are perpendicular to the magnetic field. A simulation shows, that the $B_{6-Pack}$ value can reach values of > 30 for different designs.

This new design will be more sensitive to internal fields than the so far proposed sensor. Additionally, reducing the number of turns in the windings can increase the upper bandwidth with the acceptance of a smaller signal amplitude. The current prototype showed an amplitude of several 10 volts which had to be damped anyway by the measuring electronics by means of a voltage divider for further processing.

## VI. CONCLUSION

In this paper, we presented a novel current measurement method utilizing two open inductive pickup coils for double conductors, such as power electronic modules or capacitors. This method addresses the challenges of measuring pulse currents in environments with limited

<figure><img src="image_16.png" /><figcaption>Fig. 15. Increased Measurement Sensitivity Achieved by Adjusting the Coil Turns to Align with the Magnetic Field.</figcaption></figure>

isolation distances between conductors or even in cases where there is insufficient space between the conductors. By omitting the inner part of the coil between conductors, the sensor design maintains isolation distances while still providing sufficient current measurement sensitivity.

Experimental results confirm that the proposed method accurately measures currents in double conductors while effectively suppressing interference from external switching events. Although external fields are still detected, their influence is significantly reduced due to the increased distance between the coils and the external sources. The measurements match well with those from commercial systems, validating the effectiveness of this approach. With a slightly decreased bandwidth of the proposed solution with its 12.5 MHz compared to commercial equipment, it is still sufficient to detect states and amplitudes of the current. The manufacturing of the sensor is cheap and can be integrated into the Gate Drive Unit and the needed sensing circuitry quite simple.

In conclusion, this type of sensor offers a cost-effective and easily integrable solution to measure pulse currents in power electronics application.

Future work will focus on further optimizing the sensor design to enhance sensitivity to internal fields while reducing external interference. This includes refining the coil geometry and adjusting the number of turns to improve bandwidth while maintaining signal integrity. Additionally, integrating the sensor directly into gate driver units could enable real-time current monitoring and fault detection, further increasing the reliability of power electronic systems. Experimental validation in practical converter applications should be the next step.
REFERENCES

[1] Stefan Hain et al. "The effect of different stray inductances on the performance of various types of IGBTs — Is less always better?" In: 2015 17th European Conference on Power Electronics and Applications (EPE'15 ECCE-Europe). Sept. 2015, pp. 1-9.

[2] Álvar Mayor et al. "Commutation Behavior Analysis of a Dual 3L-ANPC-VSC Phase-Leg PEBB Using 4.5-kV and 1.5-kA HV-IGBT Modules". In: *IEEE Transactions on Power Electronics* 34.2 (Feb. 2019), pp. 1125-1141.

[3] LEM International SA. *Integrated Current Sensor FHS AH 600*. datasheet Version 0. Route du Nant-d'Avril, 1521217 Meyrin, June 2023.

[4] Klaus Vogel et al. "System Cost Reduction with Integration of Shunts in Power Modules in the Power Range above 75 kW. International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy and Energy Management". In: PCIM. [Frankfurt am Main]: VDE, June 2018, pp. 182-188.

[5] Shreyas Kulkarni and Deepak Divan. "An Edge-Intelligent, Clip-on Rogowski Current Sensor With Wide Dynamic Range". In: *IEEE Sensors Journal* 21 (2 2021), pp. 1059-1071.

[6] Slavko Mocevic et al. "Phase Current Sensor and Short-Circuit Detection based on Rogowski Coils Integrated on Gate Driver for 1.2 kV SiC MOSFET Half-Bridge Module". In: *PCIM Europe*. Portland, OR, USA. IEEE, 2018, pp. 393-400.

[7] Slavko Mocevic et al. "Phase Current Reconstruction based on Rogowski Coils Integrated on Gate Driver of SiC MOSFET Half-Bridge Module for Continuous and Discontinuous PWM Inverter Applications". In: Anaheim, CA, USA. Anaheim, CA, USA: IEEE, 2019, pp. 1029-1036.

[8] Yunfei Xu et al. "A Novel Short-Circuit Protect Method for SiC MOSFET Module Based on Simplified Rogowski Coil Current Sensor". In: Xiamen, China. Xiamen, China: IEEE, 2019, pp. 1-8.

[9] Marco Stecca et al. "Gate Driver Design for 1.2 kV SiC Module with PCB Integrated Rogowski Coil Protection Circuit". In: Vancouver, BC, Canada. Vancouver, BC, Canada: IEEE, 2021, pp. 5723-5728.

[10] Matthias Spieler et al. "Design of a High-Bandwidth Compact DC-Bus Embedded Planar Rogowski Coil for SiC MOSFET Current Sensing". In: *IEEE Transactions on Power Electronics* 39.12 (Dec. 2024), pp. 16482-16497.

[11] Dr. Hans Ertl. "Vorrichtung zur Strommessung bei Leistungskondensatoren". Deutsch. AT510380(B1). Nov. 2012.

[12] Philipp Ziegler et al. "Wide Bandwidth Current Sensor for Characterization of High Current Power Semiconductor Modules". In: Ghent, Belgium. Ghent, Belgium: IEEE, 2021, pp. 1-9.

[13] A. Ahmed et al. "Design and test of a PCB Rogowski coil for very high dI/dt detection". In: 2012 15th International Power Electronics and Motion Control Conference (EPE/PEMC). Sept. 2012, DS1a.2-1-DS1a.2-4.

[14] Daniel Lexow et al. "Optimized control method for reverse conducting IGBTs". In: 2017 19th European Conference on Power Electronics and Applications (EPE'17 ECCE Europe). IEEE, Sept. 2017, P.1-P.9.