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# Model Predictive Current Control for G2V/V2G Bidirectional Operation of Onboard Charger

Hyun-Gyu Koh, Tae-Seok Kang, Irfan Sami, Yeong-Jun Choi

► To cite this version:

Hyun-Gyu Koh, Tae-Seok Kang, Irfan Sami, Yeong-Jun Choi. Model Predictive Current Control for G2V/V2G Bidirectional Operation of Onboard Charger. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0277. hal-05082327

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Submitted on 23 May 2025

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Distributed under a Creative Commons Attribution 4.0 International License
# Model Predictive Current Control for G2V/V2G Bidirectional Operation of Onboard Charger

Hyun-Gyu Koh¹, Tae-Seok Kang¹, Irfan Sami² and Yeong-Jun Choi¹, †

¹Department of Electrical Engineering, Jeju National University

102 Jejudaehak-ro, Jeju-si, Jeju-do 63243, South Korea

Tel.: +82 / (64) - 754.3677.

²Research Development Department, Milim Syscon Co. Ltd., South Korea

E-Mail: ko9003@stu.jejunu.ac.kr, wpwn1207@jejunu.ac.kr, Irfansamimwt@gmail.com,

†yeongjun.choi@jejunu.ac.kr

## Acknowledgements

This work was supported by the Industrial Strategic Technology Development Program (RS-2024-00409108, Development of Component Technology for High-Load Power Components) funded by the Ministry of Trade, Industry & Energy (MOTIE, Korea) and supported by the National Research Foundation of Korea (NRF) grant funded by the Korea government (MSIT) (RS-2024-00405278).

## Keywords

«Electric vehicle onboard charger», «Totem Pole PFC», «CLLC resonant», «Grid-to-Vehicle», «Vehicle-to-Grid», «Model Predictive Control».

## Abstract

This paper proposes a straightforward method using model predictive current control (MPCC) to achieve bidirectional power transfer in electric vehicle on-board chargers (OBCs). Unlike existing studies requiring separate controller structure changes and control gain adjustments for grid-to-vehicle (G2V) and vehicle-to-grid (V2G) modes, the proposed method enables seamless mode switching. Utilizing a totem-pole power factor correction (PFC) converter, with the primary leg switching at high frequency and the secondary leg at grid frequency, and a CLLC resonant converter, the system achieves soft switching and bidirectional power transfer. The proposed strategy maintains high power factor greater than 0.99 and input current total harmonic distortion (THD) less than 3% in G2V mode, ensures stable 180-degree phase-shifted power transfer in V2G mode, and maintains stable soft switching at the resonant frequency in both modes.

## Introduction

With the global increase in the adoption of electric vehicles (EVs), interest in vehicle-to-grid (V2G) technology has been growing. Through V2G, EVs serve beyond their primary role as transportation by returning power charged via grid-to-vehicle (G2V) to the grid, thereby improving grid stability and efficiency [1], [2].

V2G technology also plays an important role in solving the intermittency problem of renewable energy sources. When solar and wind power generation exceeds the capacity, EV batteries are used as energy storage devices (ESS) to store surplus power and release it when needed, and furthermore, this power can be supplied to the electrolysis system to produce green hydrogen. Through this, V2G technology can contribute to the activation of the hydrogen economy linked to renewable energy beyond simple power grid support. In addition, when V2G is activated, stable power supply must be provided to low-voltage (12V, 48V) electronic components in EVs, so energy management for this is also important.

The bidirectional on-board charger (OBC) supporting G2V and V2G functions consists of two main components: a power factor correction (PFC) converter that makes the phase of the grid AC side voltage and current the same to make the power factor (PF) close to 1, and a DC-DC converter that converts the DC output from the grid side AC into DC with the voltage value required by the battery. First, the isolated DC-DC converter is connected to the battery and enhances power conversion efficiency by employing soft-switching techniques. In particular, CLLC resonant converters, which provide high efficiency and the same voltage gain in both forward and reverse operations, are mainly used and actively studied [3], [4].
<figure><img src="image_3.png" /><figcaption>Fig. 1: Bi-directional OBC topology demonstration.</figcaption></figure>

Meanwhile, the totem pole PFC converter is connected to the AC grid and is responsible for processing AC-DC and DC-AC bidirectional power flow. Various studies are being conducted to design a converter with high efficiency and power density. As shown in Fig. 1, the primary leg of the totem pole PFC converter uses a wide band gap (WBG) switch because it operates at a high frequency. On the other hand, the secondary leg operates at a low frequency, which is the grid frequency, so studies are being conducted to improve the power conversion efficiency by utilizing Si devices [5]-[7]. The PFC converter and the DC-DC converter are interconnected via a DC link, enabling the OBC to effectively perform G2V and V2G functionalities in electric vehicles. In particular, various studies are conducted to support bidirectional power flow by operating the totem pole PFC converter as an inverter in V2G mode [8], [9].

This approach requires changes to the controller structure to transition from PFC to inverter control and additional design efforts to select control gains suited to the new control objectives. Consequently, it increases the complexity of controller design and necessitates further design and verification processes to ensure system stability and performance. Additionally, since both the primary and secondary legs operate at high frequencies, this poses a problem that limits the effectiveness of the previously mentioned method of optimizing the switching elements of each leg to improve efficiency. However, MPCC can improve the performance of power converters by optimizing switching while eliminating the need for controller design and reducing system complexity while maintaining high control accuracy [10]-[12].

By applying MPCC, it is possible to reduce the complexity of controller structure changes and control gain selection in PFC control and inverter

control while maintaining high control accuracy. Therefore, this paper proposes a control method that combines the high-speed switching of the primary leg and the low-speed switching of the secondary leg by utilizing MPCC to effectively realize the bidirectional power transfer of OBC. The proposed control method is providing power factor correction with a simple structure even when the power flow is switched in both directions. Specifically, in G2V mode, the input voltage and input current are in-phase, and the total harmonic distortion (THD) of the input current is maintained below 3%, and in V2G mode, the input voltage and input current are designed to be 180 degrees out of phase to maintain high power factor. In addition, the CLLC resonant circuit achieves stable soft switching at the resonant frequency.

## Overview and analysis of OBC

The circuit topology of the bidirectional charging system discussed in this paper is illustrated in Fig. 1. This system is designed to facilitate bidirectional energy flow between the grid and the high-voltage battery of an electric vehicle, operating in two modes: G2V and V2G.

In G2V mode, the primary function of the system is to draw power from the AC grid, perform power factor correction, and deliver energy efficiently and reliably to the vehicle's battery. To achieve this, a totem pole PFC converter is employed at the front end. This converter ensures a high power factor while minimizing current harmonic distortion. Additionally, it regulates the DC-link voltage, providing a stable intermediate stage for subsequent power conversion processes. Following the PFC stage, a CLLC resonant converter is used to transfer energy from the DC link to the high-voltage battery of the vehicle. The resonant circuit, designed to operate near its resonant frequency, minimizes switching losses
<figure><img src="image_4.png" /><figcaption>Fig. 2: CLLC resonant converter schematic. and achieves soft switching conditions, enabling highly efficient power transfer.</figcaption></figure>

In V2G mode, the energy flow is reversed. The CLLC resonant converter converts the energy from the battery into a stable DC-link voltage while operating at or near the resonant frequency to maintain efficiency. This DC voltage is then processed by the totem-pole PFC converter, which converts it into AC power for grid integration or local loads while maintaining high power quality and synchronization with grid parameters.

## A. CLLC resonant converter

Fig. 2 shows the schematic of the CLLC resonant converter.

Each half-bridge upper and lower has a complementary switching operation and the duty ratio D is 0.5. The phase difference between $S_1$ and $S_3$, $S_5$ and $S_7$ is 180 degrees, and $S_1$ and $S_5$ are in phase. The passive elements of the resonant tank are consist of a primary resonant inductor $L_{r1}$, a secondary resonant inductor $L_{r2}$, magnetizing inductance component $L_m$, primary resonant capacitor $C_{r1}$, secondary resonant capacitor $C_{r2}$, and high-frequency transformer with a turns ratio n:1.

The resonant tank input voltage $V_{AB}$ is represented by the fundamental wave component (1) and its root mean square (RMS) value (2) according to the Fundamental Harmonic Approximation (FHA).

$$V_{AB} = \frac{4}{\pi} V_{dc} \sin(2\pi f_s t) \quad (1)$$

$$V_{AB\_rms} = \frac{2\sqrt{2}}{\pi} V_{dc} \quad (2)$$

where $V_{dc}$ is the DC-link voltage and $f_s$ is the switching frequency of the CLLC resonant converter.

The resonant tank output voltage $V_{CD}$ is expressed in terms of the fundamental wave component (3) and the rms value (4) according to the FHA.

<figure><img src="image_5.png" /><figcaption>Fig. 3: Equivalent circuit of CLLC resonant converter represented by FHA model.</figcaption></figure>

$$V_{CD} = \frac{4}{\pi} V_{out} \sin(2\pi f_s t - \varphi) \quad (3)$$

$$V_{CD\_rms} = \frac{2\sqrt{2}}{\pi} V_{out} \quad (4)$$

where $V_{out}$ is the output voltage of the OBC and $\varphi$ is the phase difference between the primary and secondary sides of the CLLC resonant converter.

The resonant tank output current $i_{CD}$ is in phase with the output voltage $V_{CD}$, expressed in (5), and the OBC output current $I_{out}$ is expressed in (6).

$$i_{CD} = \sqrt{2} i_{CD\_rms} \sin(2\pi f_s t - \varphi) \quad (5)$$

$$I_{out} = 2f_s \int_{0}^{1/2 f_s} |i_{CD}(t)| dt = \frac{2\sqrt{2}}{\pi} I_{CD\_rms} \quad (6)$$

Therefore, the equivalent output resistance $R_{ac}$ is expressed as in (7). The resulting equivalent circuit is shown in Fig. 3.

$$n^2 R_{ac} = \frac{8 V_{out}}{\pi^2 I_{out}} \quad (7)$$

where n is turns ratio.

The voltage gain M of the CLLC resonant converter by equivalent circuit is given by

$$Q = \frac{Z_o}{R_{eq}} = \frac{\pi^2 I_{out}}{8 V_{out}} \quad (8)$$

$$f_r = \frac{1}{2\pi\sqrt{L_{r1}C_{r1}}} = \frac{1}{2\pi\sqrt{L_{r2}C_{r2}}} \quad (9)$$

$$f_n = \frac{f_s}{f_r} \quad (10)$$

$$h = \frac{L_m}{L_{r1}} \quad (11)$$

$$M(Q, f_n, h) = \frac{1}{1 + \frac{1}{h}(1 - \frac{1}{f_n^2}) + jQ[\frac{1}{h} + 2]f_n - \frac{2h+2}{hf_n} + \frac{1}{hf_n^3}} \quad (12)$$
where Q is the quality factor, $f_r$ is the resonant frequency of the CLLC converter, $f_n$ is the normalized frequency, and h is the inductance factor, which is the ratio between the magnetizing inductance and the resonant inductance.

The derivation of the voltage gain for V2G mode is similar to that of G2V mode. In addition, the high-frequency transformer turns ratio n is determined by the ratio of the DC link voltage $V_{dc}$ to the OBC output voltage $V_{out}$, as shown in (13).

$$n = \frac{V_{dc}}{V_{out}}. \qquad (13)$$

Therefore, the normalized maximum DC voltage gain $M_{max}$ is determined by the input voltage and the maximum output voltage $V_{out\_max}$ as shown in (14). The normalized minimum DC voltage gain $M_{min}$ is determined by the input voltage and the minimum output voltage $V_{out\_min}$ as shown in (15).

$$M_{max} = \frac{nV_{out\_max}}{V_{dc}}. \qquad (14)$$

$$M_{min} = \frac{nV_{out\_min}}{V_{dc}}. \qquad (15)$$

## B. Totem pole PFC converter

Fig. 4 shows the circuit and operation of the totem pole PFC converter in G2V and V2G modes. A totem pole PFC converter consists of a grid voltage $V_g$, an inductor current $i_L$, and a DC link voltage $V_{dc}$. The circuit includes passive elements such as PFC inductor, DC link capacitor $C_{link}$, and switching elements $Q_1$, $Q_2$, $Q_3$ and $Q_4$.

Where $Q_1$ and $Q_2$ are switches that operate complementary at a high switching frequency, and $Q_3$ and $Q_4$ are switches that operate complementary at 60Hz, the same as the AC grid frequency.

In addition, when power is supplied from the AC side to the DC side, it is expressed as a G2V current flow, and when power is supplied from the DC side to the AC side, it is expressed as a V2G current flow.

Fig. 4(a), (b) show the operation of the circuit during a half cycle when the grid voltage is positive. In this case, $Q_3$ should be completely OFF and $Q_4$ is always ON. Fig. 4(a) shows the operation with $Q_1$ turned ON, which is charging

<figure><img src="image_6.png" /></figure>

<figure><img src="image_7.png" /></figure>

<figure><img src="image_8.png" /></figure>

<figure><img src="image_9.png" /><figcaption>Fig. 4: G2V and V2G mode of circuit operation of totem pole PFC converter (a) $V_g$ positive half cycle $Q_1$ ON (b) $V_g$ positive half cycle $Q_2$ ON (c) $V_g$ negative half cycle $Q_1$ ON (d) $V_g$ negative half cycle $Q_2$ ON.</figcaption></figure>

$C_{link}$ during G2V. In V2G, $Q_1$ operates as a buck PFC switching operation. Fig. 4(b) shows $Q_2$ is turned ON, and $Q_2$ operates as boost PFC switching during G2V. In V2G, $Q_2$ operates complementary to $Q_1$ to create an inductor current path. In this case, $V_{Lg}$ becomes equal to $V_g$.

Fig. 4(c), (d) show the operation of the circuit during a half cycle when the grid voltage is negative. In this case, $Q_4$ should be completely OFF and $Q_3$, is always ON. Fig. 4(c) shows the operation with $Q_1$ ON, and $Q_1$ operates as a boost PFC switching in G2V. In V2G, $Q_1$ is turned ON to create an inductor current path.
During this operation, $V_{L_g}$ becomes equal to $V_g$. Fig. 4(d) shows the operation of $Q_2$ turned ON, and in G2V, $C_{link}$ is charging as shown in Fig. 4(a). In V2G, $Q_2$ operates in a buck PFC switching operation.

As shown in Fig. 4, (16) represents the operation depending on the switching ON/OFF of the boost PFC switches $Q_1$ and $Q_2$ for the positive and negative half cycles in G2V, respectively.

$$L_g \frac{di_L}{dt} = V_g D + (V_g - V_{dc})(1 - D). \quad (16)$$

Depending on the switching ON/OFF operation of the buck PFC switches $Q_2$ and $Q_1$ for the positive and negative half cycles, respectively, in V2G, (17) represents the operation.

$$L_g \frac{di_L}{dt} = (V_{dc} - V_g)D - V_g(1-D). \quad (17)$$

The relationship between inductor current ripple and inductance is set to an inductance value such that the maximum inductor current ripple is not exceed 20%, as in (18).

$$L_g \geq \frac{V_g T_s}{\Delta i_L} D \quad (18)$$

where $\Delta i_L$ is the inductor current ripple.

In the case of an AC-DC PFC converter, the DC-link voltage ripple component exists as a component twice the grid frequency, and the DC-link voltage ripple is set to a capacitance value that is not exceed 10V as in (19).

$$C_{link} \geq \frac{P_o}{4\pi V_{dc} \Delta V_{dc}} T_s \quad (19)$$

where $P_o$ is the rated power and $\Delta V_{dc}$ is the ripple in the DC link voltage.

## Proposed control strategy

Fig. 5 shows the control block diagram for control of a bidirectional OBC system. The mode selection allows the user to select G2V mode or V2G mode as 1 or 2. The totem pole PFC converter uses the mode selection to select G2V mode or V2G mode and change the control method accordingly.

In G2V mode, the voltage controller generates an output with zero error between the reference

<figure><img src="image_10.png" /><figcaption>Fig. 5: Control strategy of bidirectional OBC system.</figcaption></figure>

voltage and the DC link voltage, and in V2G mode, the output is generated by calculating the current for the rated power.

This is multiplied by the full-wave rectified waveform of the grid voltage phase through the PLL to generate the inductor current reference.

The PFC inductor current is then controlled using a prediction model for V2G and G2V modes.

To explain the proposed control method, we use Euler's law to express it in discrete time as in (20).

$$L_g \frac{di_L}{dt} = L_g \frac{i_L(k+1) - i_L(k)}{T_s} \quad (20)$$

where $T_s$ is the sampling time.

The prediction model of the inductor current of the next switching period in G2V mode obtained by using (20) is shown as (21) for ON and (22) for OFF according to the boost PFC switching state through (16).

$$i_{L,G2V,ON}(k+1) = i_L(k) + \frac{T_s}{L_g} |V_g| \quad (21)$$

$$i_{L,G2V,OFF}(k+1) = i_L(k) + \frac{T_s}{L_g} (|V_g| - V_{dc}) \quad (22)$$

where $i_{L,G2V}(k+1)$ is the inductor current prediction model for the next switching period in G2V mode.
The prediction model of the inductor current of the next switching period in V2G mode obtained by using (20) is shown as (23) for ON and (24) for OFF according to the buck PFC switching state through (17).

$$i_{L,V2G,ON}(k+1) = i_{L}(k) + \frac{T_{s}}{L_{g}}(V_{dc} - |V_{g}|) \quad (23)$$

$$i_{L,V2G,OFF}(k+1) = i_{L}(k) - \frac{T_{s}}{L_{g}}|V_{g}| \quad (24)$$

where $i_{L,V2G}(k+1)$ is the inductor current prediction model for the next switching period in V2G mode.

The cost function for switch ON/OFF derived from G2V mode is as follows (25), (26).

$$g_{G2V,ON} = |i_{L\_ref}(k+1) - i_{L,G2V,ON}(k+1)| \quad (25)$$

$$g_{G2V,OFF} = |i_{L\_ref}(k+1) - i_{L,G2V,OFF}(k+1)| \quad (26)$$

where $g_{G2V}$ is the G2V mode cost function and the difference between the inductor current reference $i_{L-ref}$ and the inductor current prediction model.

On the other hand, the cost function for switch ON/OFF derived from V2G mode is as follows (27), (28).

$$g_{V2G,ON} = |i_{L\_ref}(k+1) - i_{L,V2G,ON}(k+1)| \quad (27)$$

$$g_{V2G,OFF} = |i_{L\_ref}(k+1) - i_{L,V2G,OFF}(k+1)| \quad (28)$$

where $g_{V2G}$ is the V2G mode cost function and the difference between the inductor current reference $i_{L-ref}$ and the inductor current prediction model.

In conclusion, the ON/OFF cost functions $g$ of G2V and V2G mode are compared as in (29), and the optimal cost function $g_{opt}$, which is the minimum value between of the cost functions, determines the switching state.

$$g_{opt} = \begin{cases} \min(g_{G2V,ON}, g_{G2V,OFF}), & \text{G2V mode} \\ \min(g_{V2G,ON}, g_{V2G,OFF}), & \text{V2G mode} \end{cases} \quad (29)$$

In the control block diagram of the CLLC resonant converter in Fig. 5, the G2V mode enables PI control and pulse frequency modulation (PFM). Therefore, the duty ratio of each switch is fixed at 0.5 and the switching frequency regulates the OBC output voltage to the desired value. In V2G mode, the DC link voltage

<figure><img src="image_11.png" /><figcaption>Fig. 6: Voltage gain curves of the CLLC resonant converter in G2V and V2G modes</figcaption></figure>

is regulated by the same control method as in G2V.

Fig. 6 shows the gain curve for switching frequency $f_s$ using the voltage gain expression (12) for the CLLC resonant converter. It shows that the PFM is used to operate over a wide output voltage range for a bidirectional charging system. When operating at high output voltage conditions, the switching frequency operates in a region below the resonant frequency. Define this region as region 1 and perform boost operation. When operating under low output voltage conditions, the switching frequency operates in a region above the resonant frequency. Define this region as Region 2 and perform buck operation. Thus, the switching frequency is adjusted as the output voltage changes to regulate the output voltage to the desired value.

## Verification result

To validate the proposed control strategy, MATLAB based PLECS simulation is used, and the circuit parameters for the validation are shown in Table 1.

A 3.3 kW bidirectional OBC with an AC input voltage of 220Vrms/60Hz and a battery-side output voltage range of 180-420Vdc is used for verification.

Fig. 7 shows the resonant waveform of the CLLC in G2V mode at rated power.

Fig. 7(a) shows the drain-source voltage of the primary semiconductor switch $S_1$ and the current waveform through $S_1$. The drain-source voltage is scaled down by 1/10. As shown in the figure, the
<table><caption>**Table I: Design specifications**</caption><thead><tr><td>Symbol</td><td>Parameter</td><td>Parameter</td></tr></thead><tbody><tr><td>P<sub>o</sub></td><td>Rated power</td><td>3,300 W</td></tr><tr><td>V<sub>g</sub></td><td>Input voltage</td><td>220 Vrms</td></tr><tr><td>f<sub>g</sub></td><td>Grid frequency</td><td>60 Hz</td></tr><tr><td>V<sub>dc</sub></td><td>DC-link voltage</td><td>380 Vdc</td></tr><tr><td>L<sub>g</sub></td><td>PFC inductance</td><td>2 mH</td></tr><tr><td>C<sub>nk</sub></td><td>DC-link capacitance</td><td>1,100 μF</td></tr><tr><td>T<sub>s</sub></td><td>Sampling time</td><td>10 μs</td></tr><tr><td>f<sub>r</sub></td><td>Resonant frequency</td><td>100 kHz</td></tr><tr><td>L<sub>r1</sub>, L<sub>r2</sub></td><td>Resonant inductance</td><td>35μH</td></tr><tr><td>C<sub>r1</sub>, C<sub>r2</sub></td><td>Resonant capacitance</td><td>72 nF</td></tr><tr><td>L<sub>m</sub></td><td>Magnetizing inductance</td><td>175 μH</td></tr><tr><td>n</td><td>Turns ratio</td><td>1</td></tr><tr><td>V<sub>out</sub></td><td>Output voltage</td><td>180-420 Vdc</td></tr></tbody></table>

negative current shows anti-parallel body diode conduction across the switch, indicating that when the voltage across switch $S_1$ drops to zero, it subsequently performs a switching action, achieving zero voltage switching (ZVS).

Fig. 7(b) shows the waveforms of resonant and magnetizing currents when the DC link voltage is 380Vdc and the battery side output voltage is 420Vdc. As shown in the figure, it operates in the below region, where the switching frequency is lower than the resonant frequency $f_r$.

Fig. 8 shows the resonant waveform of CLLC in V2G mode at rated power.

Fig. 8(a) shows the drain-source voltage of the secondary-side semiconductor switch $S_5$ and the current waveform through $S_5$. The drain-source voltage is scaled down by 1/10. ZVS is achieved as shown in Fig. 7(a).

Fig. 8(b) shows the waveforms of the resonant and magnetizing currents when the battery side output voltage is 420Vdc and the DC link voltage is 380Vdc. As shown in the figure, the switching frequency is higher than the resonant frequency $f_r$ and operates in the above region.

Fig. 9 shows the G2V mode PFC waveforms at rated power.

Fig. 9(a) is the current waveform between the totem pole PFC and the DC-link capacitor. It is shown the power flow from the AC grid side to the DC-link side.

Fig. 9(b) shows the $V_{dc}$ scaled down by 1/10, the grid voltage $V_g$ scaled down by 1/10, and the

<figure><img src="image_12.png" /><figcaption>Fig. 7: CLLC resonant waveform in G2V mode of bidirectional OBC. (a) Drain-source voltage and current of $S_1$ (b) Resonant current and magnetizing current.</figcaption></figure>

<figure><img src="image_13.png" /><figcaption>Fig. 8: CLLC resonant waveform in V2G mode of bidirectional OBC. (a) Drain-source voltage and current of $S_5$ (b) Resonant current and magnetizing current.</figcaption></figure>

grid-side inductor current waveform. It is shown that the DC-link voltage is stable at 380Vdc, and the inductor current is also well controlled. As a result, the THD of the inductor current is less than
<figure><img src="image_14.png" /></figure>

<figure><img src="image_15.png" /><figcaption>Fig. 9: Totem pole PFC waveform in G2V mode of bidirectional OBC. (a) DC-link current (b) DC-link voltage $V_{dc}$, Grid voltage $V_g$ and inductor current $i_L$.</figcaption></figure>

<figure><img src="image_16.png" /></figure>

<figure><img src="image_17.png" /><figcaption>Fig. 10: Totem pole PFC waveform in V2G mode of bidirectional OBC. (a) DC-link current (b) DC-link voltage $V_{dc}$, Grid voltage $V_g$ and inductor current $i_L$.</figcaption></figure>

3%, and a high power factor of more than 0.99 is maintained in-phase with the grid voltage.

Fig. 10 shows the V2G mode PFC waveforms at rated power.

Fig. 10(a) is the same current waveform between the totem pole PFC and the DC-link capacitor as in Fig. 8(a). It is shown that the power flows from the DC-link side to the AC grid side. The DC-link voltage is stably controlled at 380 Vdc and the inductor current is also well controlled as a sinusoidal current with a phase difference of 180 degrees with the grid voltage.

## Conclusion

This paper proposes a straightforward bidirectional power transfer method for OBCs for electric vehicles using MPCC. Existing OBC bidirectional power transfer studies focus on control of a totem pole PFC converter for AC-DC conversion and DC-AC conversion, respectively, or utilizing a totem-pole PFC converter as a single-phase inverter. These existing approaches suffer from high controller design complexity. In this paper, a bidirectional power transfer control method for G2V and V2G modes based on MPCC is presented, which is based on switching only the primary leg of a totem pole PFC at high frequency. MATLAB based PLECS simulation is used to validate the proposed method, and the results show that the CLLC resonant circuit realizes soft

switching according to the resonant frequency, and the totem pole PFC circuit achieves a high power factor of more than 0.99 and THD less than 3% through the inductor current in G2V mode, and the reverse power transfer is achieved by the phase difference between the input voltage and input current in V2G mode.

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